Radio Handbook 16 1962
User Manual: Radio-Handbook-16-1962
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1.1161ffliale:n0 This book is revised and brought up to date (at irregular intervals) os necessitated by technical progress. 111E 11111110 II 1111ß001i Sixteenth Edition WILLIAM I. ORR, W6SAI Editor, 16th Edition The Standard of the Field for - advanced amateurs practical radiomen practical engineers practical technicians Published and distributed to the electronics trade by EDITORS and ENGINEERS, Ltd. Dealers: Electronic distributors, order from us. Bookstores, Taylor, Hillside, N.J. Export (exc. Canada). order from N.M. Summerland www.americanradiohistory.com , California newsdealers order from Baker li Snyder Co., 440 Park Ave. So., N.Y. 16. libraria. THE HANDBOOK RADIO SIXTEENTH EDITION Copyright, 1962, by Editors and Engineers, Ltd. Summerland, California, U.S.A. Copyright under Pan -American Convention All Translation Rights Reserved Printed in U.S.A. The "Radio Handbook" is also available on special order in Spanish and Italian editions; French, German, and Flemish -Dutch editions are in preparation or planned. Outside North America, if more convenient, write: (Spanish) Marcombo, S.A., Av. Jose Antonio, 584, Barcelona, Spain; (Italian) Edizione C.E.L.I., Via Gandino 1, Bologna, Italy; (French, German, Flemish- Dutch) P. H. Brans, Ltd., 28 Prins Leopold St., Borgerhout, Antwerp, Belgium. Other Outstanding Books from the Same Publisher (See Announcements at Back of Book) THE RADIOTELEPHONE LICENSE MANUAL THE SURPLUS RADIO CONVERSION MANUALS THE SURPLUS HANDBOOK THE WORLD'S RADIO TUBES ( RADIO TUBE VADE MECUM) TILE WORLD'S EQUIVALENT TUBES ( EQUIVALENT TUBE VADE MECUM) THE WORLD'S TELEVISION TUBES (TELEVISION TUBE VADE MECUM) www.americanradiohistory.com THE RADIO HANDBOOK 16th Edition Table of Contents Chapter One. INTRODUCTION TO RADIO Amateur Radio -1 Station and Operator Li -2 The Amateur Bands -3 Starting Your Study -4 11 Chapter Two. DIRECT CURRENT CIRCUITS The Atom 2 -1 Fundamental Electrical Units and Relationships 2 -2 Capacitors Electrostatics 2 -3 Magnetism and Electromagnetism 2 -4 RC and RL Transients 2 -5 21 Chapter Three. ALTERNATING CURRENT CIRCUITS Alternating Current 3 -1 41 41 1 1 1 1 - 3 -2 3 -3 3 -4 3 -5 11 12 12 14 21 22 30 35 38 Resonant Circuits Nonsinusoidal Waves and Transients 53 Transformers Electric Filters 61 63 Chapter Four. VACUUM TUBE PRINCIPLES Thermionic Emission 4 -1 4 -2 The Diode 4 -3 The Triode 4 -4 Tetrode or Screen Grid Tubes Mixer and Converter Tubes 4 -5 Electron Tubes at Very High Frequencies 4 -6 4 -7 Special Microwave Electron Tubes The Cathode -Ray Tube 4 -8 4 -9 Gas Tubes 4 -10 Miscellaneous Tube Types 58 67 67 71 72 77 79 80 81 84 87 88 Chapter Five. TRANSISTORS AND SEMI -CONDUCTORS Atomic Structure of Germanium and Silicon 5 -1 Mechanism of Conduction 5 -2 The Transistor 5 -3 Transistor Characteristics 5 -4 Transistor Circuitry 5 -5 Transistor Circuits 5 -6 3 90 90 90 92 94 96 103 Chapter Six. VACUUM TUBE AMPLIFIERS 6 -1 Vacuum Tube Parameters 6 -2 Classes and Types of Vacuum -Tube Amplifiers 6 -3 6 -8 Biasing Methods Distortion in Amplifiers Resistance- Capacitance Coupled Audio- Frequency Amplifiers Video -Frequency Amplifiers Other Interstage Coupling Methods Phase Inverters 6 -9 D -C 6 -10 Single -ended Triode Amplifiers Single -ended Pentode Amplifiers Push -Pull Audio Amplifiers Class B Audio Frequency Power Amplifiers Cathode- Follower Power Amplifiers Feedback Amplifiers Vacuum -Tube Voltmeters 6 -4 6 -5 6 -6 6 -7 6 -11 6 -12 6 -13 6 -14 6 -15 6 -16 Amplifiers 106 106 107 108 109 109 113 113 115 117 118 120 121 123 127 129 130 Chapter Seven. HIGH FIDELITY TECHNIQUES 7 -1 The Nature of Sound 7 -2 The Phonograph 7 -3 The High Fidelity Amplifier 7 -4 Amplifier Construction 7 -5 The "Baby Hi Fi" 7 -6 A Transformerless 25 Watt Music Amplifier 134 134 136 138 142 143 146 Chapter Eight. RADIO FREQUENCY VACUUM TUBE AMPLIFIERS Tuned RF Vacuum Tube Amplifiers 8 -1 Grid Circuit Considerations 8 -2 Plate- Circuit Considerations Radio- Frequency Power Amplifiers 8 -3 Class C. R -F Power Amplifiers 8 -4 Class B Radio Frequency Power Amplifiers 8 -5 Special R -F Power Amplifier Circuits 8 -6 Class ABI Radio Frequency Power Amplifiers 151 151 151 153 154 154 159 162 166 Chapter Nine. THE OSCILLOSCOPE 9 -1 A Typical Cathode -Ray Oscilloscope Display of Waveforms 9 -2 9 -3 Lissajous Figures 9 -4 Monitoring Transmitter Performance with the Oscilloscope 9 -5 Receiver I -F Alignment with an Oscilloscope 9 -6 Single Sideband Applications 170 Chapter Ten. SPECIAL VACUUM TUBE CIRCUITS 10 -1 Limiting Circuits 10 -2 Clamping Circuits 10 -3 Multivibrators 10 -4 The Blocking Oscillator 10 -5 Counting Circuits 10 -6 Resistance - Capacity Oscillators 10 -7 Feedback 185 185 187 188 190 190 4 www.americanradiohistory.com 170 175 176 179 180 182 191 192 194 Chapter Eleven. ELECTRONIC COMPUTERS Digital Computers 11 -1 Binary Notation 11 -2 Analog Computers 11 -3 -4 -5 11 -6 11 -7 11 11 195 195 197 199 The Operational Amplifier Solving Analog Problems Non -linear Functions Digital Circuitry 200 202 204 Chapter Twelve. RADIO RECEIVER FUNDAMENTALS Detection or Demodulation 12 -1 Superregenerative Receivers 12 -2 Superheterodyne Receivers 12 -3 Mixer Noise and Images 12 -4 211 Stages 12 -5 R -F 12 -6 Signal- Frequency Tuned Circuits I -F Tuned Circuits Detector, Audio, and Control Circuits 12 -7 12 -8 12 -9 12 -10 12 -11 12 -12 Noise Suppression Special Considerations in Receiver Adjustment Receiving Accessories U -H -F Receiver Design Chapter Thirteen. GENERATION OF RADIO FREQUENCY ENERGY Self -Controlled Oscillators 13 -1 Quartz Crystal Oscillators 13 -2 Crystal Oscillator Circuits 13 -3 Radio Frequency Amplifiers 13 -4 Neutralization of R.F. Amplifiers 13 -5 13 -6 13 -7 13 -8 13 -9 13 -10 13 -11 13 -12 13 -13 13 -14 13 -15 Neutralizing Procedure Grounded Grid Amplifiers Frequency Multipliers Tank Circuit Capacitances L and Pi Matching Networks Grid Bias Protective Circuits for Tetrode Transmitting Tubes Interstage Coupling Radio- Frequency Chokes Parallel and Push -Pull Tube Circuits Chapter Fourteen. 14 -1 14 -2 14 -3 R -F FEEDBACK Feedback Circuits Feedback and Neutralization of a Two -Stage R -F Amplifier Neutralization Procedure in Feedback -Type Amplifiers R-F Chapter Fifteen. AMPLITUDE MODULATION .. .. Sidebands .. 15 -1 Mechanics of Modulation 15 -2 Systems of Amplitude Modulation 15 -3 15 -4 15 -5 205 205 207 208 210 Input Modulation Systems Cathode Modulation 5 www.americanradiohistory.com 214 216 223 225 229 233 234 237 237 242 245 249 250 253 256 256 259 263 265 267 268 270 271 272 272 275 277 280 280 281 283 290 295 15 -6 -7 15 -8 15 The Doherty and the Terman- Woodyard Modulated Amplifiers 296 298 The Bias -Shift Heising Modulator 305 Speech Clipping Chapter Sixteen. FREQUENCY MODULATION AND RADIOTELETYPE TRANSMISSION 16 -1 Frequency Modulation 16 -2 Direct FM Circuits 16 -3 Phase Modulation 16 -4 Reception of FM Signals 16 -5 Radio Teletype Chapter Seventeen. SIDEBAND TRANSMISSION 17 -1 Commercial Applications of SSB 17 -2 Derivation of Single -Sideband Signals 17 -3 Carrier Elimination Circuits 17 -4 Generation of Single -Sideband Signals 17 -5 Single Sideband Frequency Conversion Systems 17 -6 Distortion Products Due to Nonlinearity of R-F Amplifiers 17 -7 Sideband Exciters 17 -8 Reception of Single Sideband Signals 17 -9 Double Sideband Transmission 17 -10 The Beam Deflection Modulator Chapter Eighteen. TRANSMITTER DESIGN 18 -1 Resistors 18 -2 Capacitors Wire and Inductors Grounds Holes, Leads and Shafts Parasitic Resonances Parasitic Oscillation in R-F Amplifiers Elimination of V -H -F Parasitic Oscillations Checking for Parasitic Oscillations 18 -3 18 -4 18 -5 18 -6 18 -7 18 -8 18 -9 Chapter Nineteen. TELEVISION AND BROADCAST INTERFERENCE 19 -1 Types of Television Interference 19 -2 Harmonic Radiation 19 -3 19 -4 19 -5 Low-Pass Filters Broadcast Interference HI -FI Interference Chapter Twenty. TRANSMITTER KEYING AND CONTROL 20-1 Power Systems 20 -2 Transmitter Control Methods 20 -3 Safety Precautions 20 -4 Transmitter Keying 20 -5 Cathode Keying 20 -6 Grid Circuit Keying 20 -7 Screen Grid Keying 20 -8 Differential Keying Circuits 6 www.americanradiohistory.com 308 308 311 315 317 322 323 323 324 328 330 336 340 342 347 349 350 352 352 354 356 358 358 360 361 362 364 367 367 369 372 375 382 383 383 387 389 391 393 394 395 396 Chapter Twenty -One. RADIATION, PROPAGATION AND TRANSMISSION LINES 21 -1 21 -2 21 -3 399 .._ 399 Radiation from an Antenna General Characteristics of Antennas .- 400 Radiation Resistance and Feed -Point Impedance Antenna Directivity 403 409 21 -8 Bandwidth Propagation of Radio Waves Ground -Wave Communication Ionospheric Propagation -_ 21 -9 Transmission Lines 416 21 -10 Non -Resonant Transmission Lines 417 21 -11 Tuned or Resonant Lines 420 Line Discontinuities 421 21 -4 21 -5 21 -6 21 -7 21 -12 406 409 410 412 422 Chapter Twenty -Two. ANTENNAS AND ANTENNA MATCHING End -Fed Half -Wave Horizontal Antennas 22 -1 Center -Fed Half -Wave Horizontal Antennas 22 -2 422 423 __ 22 -3 The Half -Wave Vertical Antenna 426 22 -4 The Ground Plane Antenna 427 22 -5 The Marconi 22 -6 Space- Conserving Antennas 430 22 -7 Multi -Band Antennas Matching Non -Resonant Lines to the Antenna Antenna Construction Coupling to the Antenna System 432 Antenna Couplers A Single -Wire Antenna Tuner 450 22 -8 22 -9 22 -10 22 -11 22 -12 428 Antenna 438 444 447 452 455 Chapter Twenty-Three. HIGH FREQUENCY ANTENNA ARRAYS Directive Antennas 23 -1 Long Wire Radiators 23 -2 455 457 458 460 23 -3 The V Antenna 23 -4 The Rhombic Antenna 23 -5 Stacked -Dipole Arrays 461 23 -6 Broadside Arrays 464 23 -7 End -Fire Directivity 469 23 -8 Combination End -Fire and Broadside Arrays 471 473 473 Chapter Twenty -Four. V -H -F AND U -H -F ANTENNAS Antenna Requirements 24 -1 24 -2 Simple Horizontally- Polarized Antennas 475 24 -3 Simple Vertical -Polarized Antennas 24 -4 The Discone Antenna 24 -5 Helical Beam Antennas 476 477 479 24 -6 The Corner -Reflector and Horn -Type Antennas 481 24 -7 VHF Horizontal Rhombic Antenna Multi- Element V-H -F Beam Antennas 482 24 -8 _..... 7 www.americanradiohistory.com 484 Chapter Twenty-Five. ROTARY BEAMS 25 -1 Unidirectional Parasitic End -Fire Arrays (Yogi Type) 25 -2 The Two Element 25 -3 The Three -Element 25 -4 Feed Systems Beam .___ 490 490 490 492 _ Array 494 25 -6 for Parasitic (Yogi) Arrays Unidirectional Driven Arrays Bi- Directional Rotatable Arrays 25 -7 Construction of Rotatable Arrays 502 25 -8 Tuning the Array 505 25 -9 Antenna Rotation Systems Indication of Direction "Three- Band" Beams 509 25 -5 25 -10 25 -11 26 -3 26 -4 26 -5 501 510 510 Chapter Twenty -Six. MOBILE EQUIPMENT DESIGN AND 26 -1 Mobile Reception 26 -2 500 INSTALLATION Mobile Transmitters Antennas for Mobile Work Construction and Installation of Mobile Equipment Vehicular Noise Suppression Chapter Twenty-Seven. RECEIVERS AND TRANSCEIVERS 27 -1 Circuitry and Components 27 -2 A Simple Transistorized Portable B -C Receiver 27 -3 27 -4 An Inexpensive Bandpass-Filter Receiver A Compact Transceiver for 10 and 15 Meters 27 -6 "Siamese" Converter for Six and Two Meters A Deluxe Mobile Transceiver 27 -7 A Deluxe Receiver for the DX Operator 27 -5 Chapter Twenty- Eight. LOW POWER TRANSMITTERS AND EXCITERS 511 511 517 518 520 523 526 529 529 530 539 547 555 564 .... 577 A Transistorized 50 Mc. Transmitter and Power Supply A Deluxe 200 -Watt Tabletop Transmitter 578 Strip -Line Amplifiers for VHF Circuits A "9T0" Electronic Key 595 597 Chapter Twenty -Nine. HIGH FREQUENCY POWER AMPLIFIERS 602 602 28 -1 28 -2 28 -3 28 -4 29 -1 Power Amplifier Design 29 -2 Push -Pull Triode 29 -3 Push -Pull Tetrode 29 -4 29 -5 29 -6 29 -7 29 -8 29 -9 29 -10 29 -11 29 -12 29 -13 Amplifiers Amplifiers Tetrode Pi- Network Amplifiers Grounded -Grid Amplifier Design A 350 Watt P.E.P. Grounded -Grid Amplifier The "Tri-Bander" Linear Amplifier for 20 -15 -10 An 813 Grounded -Grid Linear Amplifier The KW -2. An Economy Grounded -Grid Linear Amplifier A Pi- Network Amplifier for C -W, A -M, or SSB Kilowatt Amplifier for Linear or Class C Operation A 2- Kilowatt P.E.P. All -Band Amplifier A 3 -1000Z Linear Amplifier 8 www.americanradiohistory.com 581 604 606 609 612 617 622 627 634 643 649 654 661 Chapter Thirty. SPEECH AND AMPLITUDE MODULATION EQUIPMENT 669 30 -1 Modulation 669 30 -2 Design of Speech Amplifiers and Modulators 672 Modulator 673 30 -3 General Purpose Triode Class 30 -4 A 10 -Watt Amplifier- Driver 30 -5 A 15 -Watt Clipper- Amplifier 677 678 30 -6 A 200 -Watt 811 -A De -Luxe Modulator 679 30 -7 Zero Bias Tetrode Modulators 683 B 684 Chapter Thirty -One. POWER SUPPLIES Power Supply Requirements 31 -1 684 -2 Rectification Circuits 689 -3 Standard Power Supply Circuits 690 31 -4 Selenium and Silicon Rectifiers 695 31 -5 100 Watt Mobile Power Supply 31 -6 31 -7 Transistorized Power Supplies Two Transistorized Mobile Supplies 697 703 706 31 -8 Power Supply Components 31 -9 Special Power Supplies 31 -10 Power Supply Design 31 -11 -_ 300 Volt, 50 Ma. Power Supply 1500 Volt, 425 Milliampere Power Supply 716 A Dual Voltage Transmitter Supply 718 718 31 31 31 -12 31 -13 -14 31 707 709 713 . . _ A Kilowatt Power Supply 717 720 Chapter Thirty -Two. WORKSHOP PRACTICE 720 723 32 -1 Tools 32 -2 The 32 -3 TVI -Proof Enclosures 724 32 -4 Enclosure Openings 32 -5 Summation of the Problem 725 725 32 -6 Construction Practice 32 -7 Shop Layout Material 726 729 731 Chapter Thirty- Three. ELECTRONIC TEST EQUIPMENT Voltage, Current and Power 33 -1 Measurement of Circuit Constants _ 33 -2 731 737 33 -3 Measurements with a Bridge 738 33 -4 Frequency Measurements 739 33 -5 Antenna and Transmission Line Measurements 740 33 -6 A Simple Coaxial 742 33 -7 Measurements on Balanced Transmission Lines 33 -8 A 33-9 The Antennascope 33 -10 A Silicon Crystal Noise Generator 747 749 33 -11 A Monitor Scope for AM and SSB 750 "Balanced" Reflectometer Chapter Thirty -Four. RADIO MATHEMATICS AND 744 745 SWR Bridge CALCULATIONS 9 www.americanradiohistory.com 752 FOREWORD TO THE SIXTEENTH EDITION Over two decades ago the historic first edition of the RADIO HANDBOOK was published as a unique, independent, communications manual written especially for the advanced radio amateur and electronic engineer. Since that early issue, great pains have been taken to keep each succeeding edition of the RADIO HANDBOOK abreast of the rapidly expanding field of electronics. So quickly has the electron invaded our everyday affairs that it is now no longer possible to segregate one particular branch of electronics and define it as radio communications; rather, the transfer of intelligence by electrical means encompasses more than the vacuum tube, the antenna, and the tuning capacitor. Included in this new, advanced Sixteenth Edition of the RADIO HANDBOOK are fresh chapters covering electronic computers, r.f. feedback amplifiers, and high fidelity techniques, plus greatly expanded chapters dealing with semi- conductors and special vacuum tube circuits. The other chapters of this Handbook have been thoroughly revised and brought up to date, touching briefly on those aspects in the industrial and military electronic fields that are of immediate interest to the electronic engineer and the radio amateur. The construction chapters have been completely re- edited. All new equipments described therein are of modern design, free of TV! producing problems and various unwanted parasitic oscillations. The writing and preparation of this Handbook would have been impossible without the lavish help that was tended the editor by fellow amateurs and sympathetic electronic organizations. Their friendly assistance and helpful suggestions were freely given in the true amateur spirit to help make the 16th edition of the RADIO HANDBOOK an outstanding success. The editor and publisher wish to thank these individuals and companies whose unselfish support made the compilation and publication of this book an interesting and inspired task. -WILLIAM I. ORR, W6SAI, 3A2AF, Editor Thomas Consalvi, W3EOZ, Barker & Williamson, Inc. Claude E. Doner, W3FAL, Radio Corporation of America John A. Evans, W9HRH, Potter & Brumfield Co. Wayne Green, W2NSD, 73 Magazine Jo Jennings, W6EI, Jennings Radio Mfg. Co. E. A. Neal, W4ITC, General Electric Co. Harold Vance, K2FF, Radio Corporation of America Blackhawk Engineering Co. H. E. Blaksley, K7ASK Byron Hunter, W6VML Clifford Johnson, WOURQ Herbert Johnson, W7GRA Thomas Lamb, K8ERV James G. Lee, W6VAT Hugh MacDonald, W6CDT Otto Miller, K6ENX Robert Moore, W7JNC B. A. Ontiveros, W6FFF (drafting) A. L. Patrick, W9EHW Raymond Rinaudo, W6KEV Robert Sutherland, W6UOV W. H. Sayer, Jr., WA6BAN Mel Whiteman, W6BZ www.americanradiohistory.com CHAPTER ONE Introduction to Radio to the teaching of the principles of equipment design and signal propagation. It is in response to requests from schools and agencies of the Department of Defense, in addition to persistent requests from the amateur radio fraternity, that coverage of these principles has been expanded. The field of radio is a division of the much larger field of electronics. Radio itself is such a broad study that it is still further broken down into a number of smaller fields of which only shortwave or high- frequency radio is covered in this book. Specifically the field of communication on frequencies from 1.8 to 450 megacycles is taken as the subject matter for this work. 1 The largest group of persons interested in the subject of high-frequency communication is the more than 350,000 radio amateurs located in nearly all countries of the world. Strictly speaking, a radio amateur is anyone interested in radio non -commercially, but the term is ordinarily applied only to those hobbyists possessing transmitting equipment and a license from the government. It was for the radio amateur, and particularly for the serious and more advanced amateur, that most of the equipment described in this book was developed. However, in each equipment group, simple items also are shown for the student or beginner. The design principles behind the equipment for high- frequency radio communication are of course the same whether the equipment is to be used for commercial, military, or amateur purposes, the principal differences lying -1 Amateur Radio Amateur radio is a fascinating hobby with many phases. So strong is the fascination offered by this hobby that many executives, engineers, and military and commercial operators enjoy amateur radio as an avocation even though they are also engaged in the radio field commercially. It captures and holds the interest of many people in all walks of life, and in all countries of the world where amateur activities are permitted by law. Amateurs have rendered much public service through furnishing communications to and from the outside world in cases where disaster has isolated an area by severing all wire com- munications. Amateurs have a proud record of heroism and service in such occasion. Many expeditions to remote places have been kept in touch with home by communication with amateur stations on the high frequencies. The amateur's fine record of performance with the "wireless" equipment of World War I has been surpassed by his outstanding service in World in construction practices, and in the tolerances and safety factors placed upon components. With the increasing complexity of high-frequency communication, resulting primarily from increased utilization of the available spectrum, it becomes necessary to delve more deeply into the basic principles underlying radio War II. By the time peace came in the Pacific in the summer of 1945, many thousand amateur operators were serving in the allied armed forces. They had supplied the army, navy, marines, coast guard, merchant marine, civil service, war plants, and civilian defense organizations with trained personnel for radio, communication, both from the standpoint of equipment design and operation and from the standpoint of signal propagation. Hence, it will be found that this edition of the RADIO HANDBOOK has been devoted in greater proportion 11 www.americanradiohistory.com Introduction to Radio 12 radar, wire, and visual communications and for teaching. Even now, at the time of this writing, amateurs are being called back into the expanded defense forces, are returning to defense plants where their skills are critically needed, and are being organized into communication units as an adjunct to civil defense groups. 1 Station and Operator Licenses -2 Every radio transmitting station in the United States no matter how low its power must have a license from the federal government before being operated; some classes of stations must have a permit from the government even before being constructed. And every operator of a transmitting station must have an operator's license before operating a transmitter. There are no exceptions. Similar laws apply in practically every major country. There are at present six classes of amateur operator licenses which have been authorized by the Federal Communications Commission. These classes differ in "Classes of Amateur Operator Li many respects, so each will be discussed briefly. (a) Amateur Extra Class. This class of license is available to any U. S. citizen who at any time has held for a period of two years or more a valid amateur license, issued by the FCC, excluding licenses of the Novice and Technician Classes. The examination for the license includes a code test at 20 words per minute, the usual tests covering basic amateur practice and general amateur regulations, and an additional test on advanced amateur practice. All amateur privileges are accorded the holders of this operator's license. (b) General Class. This class of amateur license is equivalent to the old Amateur Class B license, and accords to the holders all amateur privileges except those which may be set aside for holders of the Amateur Extra Class license. This class of amateur operator's license is available to any U. S. citizen. The examination for the license includes a code test at 13 words per minute, and the usual examinations covering basic amateur practice and general amateur regulations. (c) Conditional Class. This class of amateur license and the privileges accorded by it are equivalent to the General Class license. However, the license can be issued only to those whose residence is more than 125 miles airline from the nearest location at which FCC examinations are held at intervals of not more than three months for the General Class amateur operator license, or to those who for any THE RADIO of several specified reasons are unable to appear for examination. (d) Technician Class. This is a new class of license which is available to any citizen of the United States. The examination is the same as that for the General Class license, except that the code test is at a speed of 5 words per minute. The holder of a Technician class license is accorded all authorized amateur privileges in the amateur frequency bands above 220 megacycles, and in the 50-Mc. band. (e) Novice (.lass. this is a new class of license which is available to any U. S. citizen who has not previously held an amateur license of any class issued by any agency of the U. S. government, military or civilian. The examination consists of a code test at a speed of 5 words per minute, plus an examination on the rules and regulations essential to beginner's operation, including sufficient elementary radio theory for the understanding of those rules. The Novice Class of license affords severely restricted privileges, is valid for only a period of one year (as contrasted to all other classes of amateur licenses which run for a term of five years), and is not renewable. All Novice and Technician class examinations are given by volunteer examiners, as regular examinations for these two classes are not given in FCC offices. Amateur radio clubs in the larger cities have established examin ing committees to assist would -be amateurs of the area in obtaining their Novice and Technician licenses. 1 -3 The Amateur Bands Certain small segments of the radio frequen- cy spectrum between 1500 kc. and 10,000 .fc. are reserved for operation of amateur radio stations. These segments are in general agreement throughout the world, although certain parts of different amateur bands may be used for other purposes in various geographic regions. In particular, the 40 -meter amateur band is used legally (and illegally) for short wave broadcasting by many countries in Europe, Africa and Asia. Parts of the 80 -meter band are used for short distance marine work in Europe, and for broadcasting in South America. The amateur bands available to American radio amateurs aree The 160 -meter band is divided into 25- kilocycle segments on a regional basis, with day and night power limitations, and is available for amateur use provided no interference is caused to the Loran (Long Range Navigation) stations operating in this band. This band is least affected by the 11- 160 Meters (1800 Kc. -2000 Kc.) www.americanradiohistory.com Amateur Bands HANDBOOK year solar sunspot cycle. The Maximum Usable Frequency (MUF) even during the years of decreased sunspot activity does not usually drop below 4 Mc., therefore this band is not subject to the violent fluctuations found on the higher frequency bands. DX contacts on on this band are limited by the ionospheric absorption of radio signals, which is quite high. During winter nighttime hours the absorption is often of a low enough value to permit trans -oceanic contacts on this band. On rare occasions, contacts up to 10,000 miles have been made. As a usual rule, however, 160 -meter amateur operation is confined to ground -wave contacts or single -skip contacts of 1000 miles or less. Popular before World War II, the 160 -meter band is now only sparsely occupied since many areas of the country are blanketed by the megawatt pulses of the Loran chains. The 80 -meter band is the most popular amateur band in the continental United States for local "rag- chewing" and traffic nets. During the years of minimum sunspot activity the ionospheric absorption on this band may be quite low, and long distance DX contacts are possible during the winter night hours. Daytime operation, in general, is limited to contacts of 500 miles or less. During the summer months, local static and high ionospheric absorption limit long distance contacts on this band. As the sunspot cycle advances and the MUF rises, increased ionospheric absorption will tend to degrade the long distance possibilities of this band. At the peak of the sunspot cycle, the 80 -meter band becomes useful only for short-haul communication. 80 Meters (3500 Kc. -4000 Kc.) The 40 -meter band is high Kc) enough in frequency to be severely affected by the 11 -year sunspot cycle. During years of minimum solar activity, the MUF may drop below 7 Mc., and the band will become very erratic, with signals dropping completely out during the night hours. Ionospheric absorption of signals is not as large a problem on this band as it is on 80 and 160 meters. As the MUF gradually rises, the skip- distance will increase on 40 meters, especially during the winter months. At the peak of the solar cycle, the daylight skip distance on 40 meters will be quite long, and stations within a distance of 500 miles or so of each other will not be able to hold communication. DX operation on the 40 -meter band is considerably hampered by broadcasting stations, propaganda stations, and jamming trans40 Meters (7000 Kc. -7300 13 mitters. In Europe and Asia the band is in a chaotic state, and amateur operation in this region is severely hampered. At the present time, 20 Meters (14,000 Kc.-14,350 Kc.) the 20 -meter band is by far the most popular band for long distance contacts. High enough in frequency to be almost obliterated at the bottom of the solar cycle, the band nevertheless provides good DX contacts during years of minimal sunspot activity. At the present time, the band is open to almost all parts of the world at some time during the year. During the summer months, the band is active until the late evening hours, but during the winter months the band is only good for a few hours during daylight. Extreme DX contacts are usually erratic, but the 20 -meter band is the only band available for DX operation the year around during the bottom of the DX cycle. As the sunspot count increases and the MUF rises, the 20 -meter band will become open for longer hours during the winter. The maximum skip distance increases, and DX contacts are possible over paths other than the Great Circle route. Signals can be heard the "long paths," 180 degrees opposite to the Great Circle path. During daylight hours, absorption may become apparent on the 20 -meter band, and all signals except very short skip may disappear. On the other hand, the band will be open for worldwide DX contacts all night long. The 20 -meter band is very susceptible to "fade- outs" caused by solar disturbances, and all except local signals may completely disappear for periods of a few hours to a day or so. This is a relatively new band for radio amateurs since it has only been available for amateur operation since 1952. Not too much is known about the characteristics of this band, since it has not been occupied for a full cycle of solar activity. However, it is reasonable to assume that it will have characteristics similar to both the 20 and 10 -meter amateur bands. It should have a longer skip distance than 20 meters for a given time, and sporadic -E (short -skip) should be apparent during the winter months. During a period of low sunspot activity, the MUF will rarely rise as high as 15 meters, so this band will be "dead" for a large part of the year. During the next few years, 15 -meter activity should pick up rapidly, and the band should support extremely long DX contacts. Activity on the 15 -meter band is limited in some areas, 15 Meters (21,000 Kc.- 21,450 Kc.) www.americanradiohistory.com 14 I n t r o d u c t i o n t o R a d i o since the older model TV receivers have a 21 Mc. i -f channel, which falls directly in the 15 -meter band. The interference problems brought about by such an unwise choice of intermediate frequency often restrict operation on this band by amateur stations unfortunate enough to be situated near such an obsolete receiver. Meters (28.000 Kc.- 29,700 Kc.) 10- During the peak of the sunspot cycle, the 10meter band is without doubt the most popular amateur band. The combination of long skip and low ionospheric absorption make reliable DX contacts with low powered equipment possible. The great width of the band (1700 kc.) provides room for a large number of amateurs. The long skip(1500 miles or so) prevents nearby amateurs from hearing each other, thus dropping the interference level. During the winter months, sporadic -E (short skip) signals up to 1200 miles or so will be heard. The 10meter band is poorest in the summer months, even during a sunspot maximum. Extremely long daylight skip is common on this band, and and in years of high MUF the 10 -meter band will support intercontinental DX contacts during daylight hours. The second harmonic of stations operating in the 10 -meter band falls directly into television channel 2, and the higher harmonics of 10 -meter transmitters fall into the higher TV channels. This harmonic problem seriously curtailed amateur 10 -meter operation during the late 40's. However, with the new circuit techniques and TVI precautionary measures stressed in this Handbook, 10 -meter operation should cause little or no interference to nearby television receivers of modern design. At the peak of the sunspot cycle, the MUF occasionally rises high enough to permit DX contacts up to 10,000 miles or so on 6 meters. Activity on this band during such a period is often quite high. Interest in this band wanes during a period of lesser solar activity, as contacts, as a rule, are restricted to short skip work. The proximity of the 6-meter band to television channel 2 often causes interference problems to amateurs located in areas where channel 2 is active. As the sunspot cycle increases, activity on the 6 -meter band will increase. Six Meters (50 Mc. -54 Mc.) The V -HF Bands (Two Meters and "Up ") v -h -f bands are the least affected by The the vagaries of the sunspot cycle and the Heaviside layer. Their predominant use is for reliable communication over distances of 150 miles or less. These T H E R A D I O bands are sparsely occupied in the rural sections of the United States, but are quite heavily congested in the urban areas of high popu- lation. In recent years it has been found that v -h -f signals are propagated by other means than by line -of-sight transmission. "Scatter signals," Aurora reflection, and air -mass boundary bending are responsible for v -h -f communication up to 1200 miles or so. Weather conditions will often affect long distance communication on the 2 -meter band, and all the v -h -f bands are particularly sensitive to this condition. The other v -h -f bands have had insufficient occupancy to provide a clear picture of their characteristics. In general, they behave much as does the 2 -meter band, with the weather effects becoming more pronounced on the higher frequency bands. 1 -4 Starting Your Study When you start to prepare yourself for amateur examination you will find that the cuit diagrams, tube characteristic curves, formulas appear confusing and difficult of derstanding. But after the cirand un- a few study sessions becomes sufficiently familiar with the notation of the diagrams and the basic concepts of theory and operation so that the acquisition of further knowledge becomes easier and even fascinating. As it takes a considerable time to become proficient in sending and receiving code, it is a good idea to intersperse technical study sessions with periods of code practice. Many short code practice sessions benefit one more than a small number of longer sessions. Alternating between one study and the other keeps the student from getting "stale" since each type of study serves as a sort of respite from the other. When you have practiced the code long enough you will be able to follow the gist of the slower sending stations. Many stations send very slowly when working other stations at great distances. Stations repeat their calls many times when calling other stations before contact is established, and one need not have achieved much code proficiency to make out their calls and thus determine their location. one The Code The applicant for any class of amateur operator license must be able to send and receive the Continental Code (sometimes called the International Morse Code). The speed required for the sending and receiving test may be either 5, 13, or 20 words per minute, depending upon the class of license, assuming an average of five characters to the word in each case. The sending and re- www.americanradiohistory.com Learning the Code HANDBOOK A 6 C D E . = MI N O P Q MO H I ,J K L M MED IEM T 7 NEI EMI V W =El 41 X MIMEE Y MIMED MI Z MD ME Ma S U -. ) 2 3 4 Mo =El 5 6 R F G MD 8 9 GM . =.. fm. gm, OM 4=1 1M ME, MED 4=1 IMP 0 MEANS ZERO. AND IS WRITTEN IN THIS WAY TO DISTINGUISH IT FROM THE LETTER 'O'' IT OFTEN IS TRANSMITTED INSTEAD AS ONE LONG DASH (EQUIVALENT TO 5 DOTS) 0 MI PERIOD (.) WAIT SIGN (AS) COMMA (,) DOUBLE DASH (BREAK) INTERROGATION (7) QUOTATION MARK (") ERROR (ERASE SIGN) COLON ( FRACTION BAR( /) END OF MESSAGE (AR) ) SEMICOLON END OF TRANSMISSION (SK) INTERNAT. DISTRESS SIG. (SOS) (I) PARENTHESIS 15 ( I Figure . _ MMD mo IMID e moo 1 rodio The Continental (or International Morse) Code is used for substantially all non-automatic and of SOUND, communication. DO NOT memorize from the printed page; code is a language must not be learned visually; learn by listening as explained in the text. ceiving tests run for five minutes, and one minute of errorless transmission or reception must be accomplished within the five -minute interval. If the code test is failed, the applicant must wait at least one month before he may again appear for another test. Approximately 30% of amateur applicants fail to pass the test. It should be expected that nervousness and excitement will at least to some degree temporarily lower the applicant's code ability. The best prevention against this is to master the code at a little greater than the required speed under ordinary conditions. Then if you slow down a little due to nervousness during a test the result will not prove fatal. There is no shortcut to code pro ficiency. To memorize the alphabet entails but a few evenings of diligent application, but considerable time is required to build up speed. The exact time required depends upon the individual's ability and the regularity of practice. While the speed of learning will naturally vary greatly with different individuals, about 70 hours of practice (no practice period to be over 30 minutes) will usually suffice to bring a speed of about 13 w.p.m.; 16 w.p.m. requires about 120 hours; 20 w.p.m., 175 hours. Memorizing the Code Since code reading requires that individual letters be recognized instantly, any memoriz- ing scheme which depends upon orderly se- quence, such as learning all "dab" letters and all "dit" letters in separate groups, is to be discouraged. Before beginning with a code practice set it is necessary to memorize the whole alphabet perfectly. A good plan is to study only two or three letters a day and to drill with those letters until they become part of your consciousness. Mentally translate each day's letters into their sound equivalent wherever they are seen, on signs, in papers, indoors and outdoors. Tackle two additional letters in the code chart each day, at the same time reviewing the characters already learned. Avoid memorizing by routine. Be able to sound out any letter immediately without so much as hesitating to think about the letters preceding or following the one in question. Know C, for example, apart from the sequence ABC. Skip about among all the characters learned, and before very long sufficient letters will have been acquired to enable you to spell out simple words to yourself in "dit dabs." This is interesting exercise, and for that reason it is good to memorize all the vowels first and the most common consonants next. Actual code practice should start only when the entire alphabet, the numerals, period, corn- www.americanradiohistory.com Introduction to Radio 16 THE RADIO tion, do it in code. It makes more interesting practice than confining yourself to random practice material. hen two co- learners have memorized the code and are ready to start sending to each other for practice, it is a good idea to enlist the aid of an experienced operator for the first practice session or two so that they will get an idea of how properly formed characters sound. Figure 2 These code characters are used in languages other than English. They may occasionally be encountered so it is well to know them. ma, and question mark have been memorized so thoroughly that any one can be sounded without the slightest hesitation. Do not bother with other punctuation or miscellaneous signals until later. - Each letter and figure must be memorized by its sound rather than its appearance. Code is a system of sound communication, the same as is the spoken word. The letter A, for example, is one short and one long sound in combination sounding like dit dab, and it must be remembered as such, and not as "dot dash." Sound Not Sight Practice Time, patience, and regularity are required to learn the code properly. Do not expect to accomplish it within a few days. Don't practice too long at one stretch; it does more harm than good. Thirty minutes at a time should be the limit. Lack of regularity in practice is the most common cause of lack of progress. Irregular practice is very little better than no practice at all. Write down what you have heard; then forget it; do not look back. If your mind dwells even for an instant on a signal about which you have doubt, you will miss the next few characters while your attention is diverted. While various automatic code machines, phonograph records, etc., will give you practice, by far the best practice is to obtain a study companion who is also interested in learning the code. When you have both memorized the alphabet you can start sending to each other. Practice with a key and oscillator or key and buzzer generally proves superior to all automatic equipment. Two such sets operated between two rooms are fine -or between your house and his will be just that much better. Avoid talking to your partner while practicing. If you must ask him a ques- During the first practice period the speed should be such that substantially solid copy can be made without strain. Never mind if this is only two or three words per minute. In the next period the speed should be increased slightly to a point where nearly all of the characters can be caught only through conscious effort. When the student becomes proficient at this new speed, another slight increase may be made, progressing in this manner until a speed of about 16 words per minute is attained if the object is to pass the amateur 13 -word per minute code test. The margin of 3 w.p.m. is recommended to overcome a possible excitement factor at examination time. Then when you take the test you don't have to worry about the "jitters" or an "off day." Speed should not be increased to a new level until the student finally makes solid copy with ease for at least a five -minute period at the old level. How frequently increases of speed can be made depends upon individual ability and the amount of practice. Each increase is apt to prove disconcerting, but remember "you are never learning when you are comfortable." A number of amateurs are sending code practice on the air on schedule once or twice each week; excellent practice can be obtained after you have bought or constructed your re- ceiver by taking advantage of these sessions. If you live in a medium -size or large city, the chances are that there is an amateur radio club in your vicinity which offers free code practice lessons periodically. Skill listen to someone speaking you do not consciously think how his words are spelled. This is also true when you read. In code you must train your ears to read code just as your eyes were trained in school to read printed matter. With enough practice you acquire skill, and from skill, speed. In other words, it becomes a habit, something which can be done without conscious effort. Conscious effort is fatal to speed; we can't think rapidly enough; a speed of 25 words a minute, which is a common one in commercial operations, means 125 characters per minute or more than two per second, which leaves no time for conscious thinking. When you www.americanradiohistory.com Learning the Code HANDBOOK Perfect Formation of Characters When transmitting on the code practice set to your partner, concentrate on the quality of your sending, not on your speed. Your partner will appreciate it and he could not copy you if you speeded up anyhow. If you want to get a reputation as having an excellent "fist" on the air, just remember that speed alone won't do the trick. Proper execution of your letters and spacing will make much more of an impression. Fortunately, as you get so that you can send evenly and accurately, your sending speed will automatically increase. Remember to try to see how evenly you can send, and how fast you can receive. Concentrate on making signals properly with your key. Perfect formation of characters is paramount to everything else. Make every signal right no matter if you have to practice it hundreds or thousands of times. Never allow yourself to vary the slightest from perfect formation once you have learned it. If possible, get a good operator to listen to your sending for a short time, asking him to criticize even the slightest imperfections. Timing It is of the utmost importance to maintain uniform spacing in characters and combinations of characters. Lack of uniformity at this point probably causes beginners more trouble than any other single factor. Every dot, every dash, and every space must be correctly timed. In other words, accurate timing is absolutely essential to intelligibility, and timing of the spaces between the dots and dashes is just as important as the lengths of the dots and dashes themselves. The characters are timed with the dot as a "yardstick." A standard dash is three times as long as a dot. The spacing between parts of the same letter is equal to one dot; the space between letters is equal to three dots, and that between words equal to five dots. The rule for spacing between letters and words is not strictly observed when sending slower than about 10 words per minute for the benefit of someone learning the code and desiring receiving practice. When sending at, say, 5 w.p.m., the individual letters should be made the same as if the sending rate were about 10 w.p.m., except that the spacing between letters and words is greatly exaggerated. The reason for this is obvious. The letter L, for instance, will then sound exactly the same at 10 w.p.m. as at 5 w.p.m., and when the speed is increased above 5 w.p.m. the student will not have to become familiar with what may seem to him like a new sound, although it is in reality only a faster combination of dots and dashes. At the greater speed he will merely have to learn the identification of the same sound without taking as long to do so. 17 Or-:C,t,. 0oó000boá taMS ins C B tmo tit tt> riti A O Figure IMP N E 3 Diagram illustrating relative lengths of dashes and spaces referred to the duration of o dot. A dash is exactly equal in duration to three dots; spaces between parts of a letter equal one dot; those between letters, three dots; space between words, five dots. Note that a slight increase between two parts of a letter will make it sound like two letters. Be particularly careful of letters like B. Many beginners seem to have a tendency to leave a longer space after the dash than that which they place between succeeding dots, thus making it sound like TS. Similarly, make sure that you do not leave a longer space after the first dot in the letter C than you do between other parts of the same letter; otherwise it will sound like NN. Once you have memorized the code thoroughly you should concentrate on increasing your receiving speed. True, if you have to practice with another newcomer who is learning the code with you, you will both have to do some sending. But don't attempt to practice sending just for the sake of increasing your sending speed. When transmitting on the code practice set to your partner so that he can get receiving practice, concentrate on the quality of your sending, not on your speed. Because it is comparatively easy to learn to send rapidly, especially when no particular care is given to the quality of sending, many operators who have just received their licenses get on the air and send mediocre or worse code at 20 w.p.m. when they can barely receive good code at 13. Most oldtimers remember their own period of initiation and are only too glad to be patient and considerate if you tell them that you are a newcomer. But the surest way to incur their scorn is to try to impress them with your "lightning speed," and then to request them to send more slowly when they come back at you at the same speed. Stress your copying ability; never stress your sending ability. It should be obvious that that if you try to send faster than you can receive, your ear will not recognize any mistakes which your hand may make. Sending vs. Receiving www.americanradiohistory.com 1 8 I n t ro d u c t i o n t o R a d i T o H E R A D I O fingers to become tense. Send with a full, free arm movement. Avoid like the plague any finger motion other than the slight cushioning effect mentioned above. Stick to the regular hand key for learning code. No other key is satisfactory for this purpose. Not until you have thoroughly mastered both sending and receiving at the maximum speed in which you are interested should you tackle any form of automatic or semi -automatic key such as the Vibroplex ( "bug ") or an electronic key. Difficulties Figure 4 PROPER POSITION OF THE FINGERS FOR OPERATING A TELEGRAPH KEY The fingers hold the knob and act os a cushion. The hand rests lightly on the key. The muscles of the forearm provide the power, the wrist acting as the fulcrum. The power should not come from the fingers, but rather from the forearm muscles. Figure 4 shows the proper position of the hand, fingers and wrist when manipulating a telegraph or radio key. The forearm should rest naturally on the desk. It is preferable that the key be placed far enough back from the edge of the table (about 18 inches) that the elbow can rest on the table. Otherwise, pressure of the table edge on the arm will tend to hinder the circulation of the blood and weaken the ulnar nerve at a point where it is close to the surface, which in turn will tend to increase fatigue considerably. The knob of the key is grasped lightly with the thumb along the edge; the index and third fingers rest on the top towards the front or far edge. The hand moves with a free up and down motion, the wrist acting as a fulcrum. The power must come entirely from the arm muscles. The third and index fingers will bend slightly during the sending but not because of deliberate effort to manipulate the finger muscles. Keep your finger muscles just tight enough to act as a cushion for the arm motion and let the slight movement of the fingers take care of itself. The key's spring is adjusted to the individual wrist and should be neither too stiff nor too loose. Use a moderately stiff tension at first and gradually lighten it as you become more proficient. The separation between the contacts must be the proper amount for the desired speed, being somewhat under 1/16 inch for slow speeds and slightly closer together (about 1/32 inch) for faster speeds. Avoid extremes in either direction. Do not allow the muscles of arm, wrist, or Using the Key Should you experience difficulty in increasing your code speed after you have once memorized the characters, there is no reason to become discouraged. It is more difficult for some people to learn code than for others, but there is no justification for the contention sometimes made that "some people just can't learn the code." It is not a matter of intelligence; so don't feel ashamed if you seem to experience a little more than the usual difficulty in learning code. Your reaction time may be a little slower or your coordination not so good. If this is the case, remember you can still learn the code. You may never learn to send and receive at 40 w.p.m., but you can learn sufficient speed for all non -commercial purposes and even for most commercial purposes if you have patience, and refuse to be discouraged by the fact that others seem to pick it up more rapidly. When the sending operator is sending just a bit too fast for you (the best speed for practice), you will occasionally miss a signal or a small group of them. When you do, leave a blank space; do not spend time futilely trying to recall it; dismiss it, and center attention on the next letter; otherwise you'll miss more. Do not ask the sender any questions until the transmission is finished. To prevent guessing and get equal practice on the less common letters, depart occasionally from plain language material and use a jumble of letters in which the usually less commonly used letters predominate. As mentioned before, many students put a greater space after the dash in the letter B than between other parts of the same letter so it sounds like TS. C, F, Q, V, X, Y and Z often give similar trouble. Make a list of words or arbitrary combinations in which these letters predominate and practice them, both sending and receiving until they no longer give you trouble. Stop everything e l s e and stick at them. So long as they give you trouble you are not ready for anything else. Follow the same procedure with letters which you may tend to confuse such as F and L, which are often confused by beginners. www.americanradiohistory.com HANDBOOK Figure Learning the Code 19 5 THE SIMPLEST CODE PRACTICE SET CONSISTS OF A KEY AND A INEXPENSIVE 500 OHM POTENTIOMETER VOLUME CONTROL BUZZER is adjusted to give a steady, high -pitched whine. If desired, the phones may be omitted, in which case the buzzer should be mounted firmly on a sounding board. Crystal, magnetic, or dynamic earphones may be used. Additional sets of phones should be connected in parallel, not in series. The buzzer = 1.5 TO S VOLTS OF BATTERY 1 KEY Keep at it until you always get them right without having to stop even an instant to think about it. If you do not instantly recognize the sound of any character, you have not learned it; go back and practice your alphabet further. You should never have to omit writing down every signal you hear except when the transmission is too fast for you. Write down what you hear, not what you think it should be. It is surprising how often the word which you guess will be wrong. All good operators copy several words behind, that is, while one word is being received, they are writing down or typing, say, the fourth or fifth previous word. At first this is very difficult, but after sufficient practice it will be found actually to be easier than copying close up. It also results in more accurate copy and enables the receiving operator to capitalize and Copying Behind CH-722 COLLECTOR 2= BASE 3= EMITTER REO 00T 2000 n PHONES 10K KEY 0.5 W Figure PHONES. TO 4 PAIR 6 SIMPLE TRANSISTOR CODE PRACTICE OSCILLATOR An inexpensive Raytheon CK -722 transistor requires only a single 11,2 -volt flashlight battery for power. The inductance of the earphone windings forms part of the oscillatory circuit. The pitch of the note may be changed by varying the value of the two capacitors connected across the earphones. THESE PARTS REQUIRED ONLY IF HEADPHONE OPERATION IS DESIRED punctuate copy as he goes along. It is not recommended that the beginner attempt to do this until he can send and receive accurately and with ease at a speed of at least 12 words a minute. It requires a considerable amount of training to dissociate the action of the subconscious mind from the direction of the conscious mind. It may help some in obtaining this training to write down two columns of short words. Spell the first word in the first column out loud while writing down the first word in the second column. At first this will be a bit awkward, but you will rapidly gain facility with practice. Do the same with all the words, and then re- verse columns. Next try speaking aloud the words in the one column while writing those in the other column; then reverse columns. After the foregoing can be done easily, try sending with your key the words in one column while spelling those in the other. It won't be easy at first, but it is well worth keeping after if you intend to develop any real code proficiency. Do not attempt to catch up. There is a natural tendency to close up the gap, and you must train yourself to overcome this. Next have your code companion send you a word either from a list or from straight text; do not write it down yet. Now have him send the next word; after receiving this second word, write down the first word. After receiving the third word, write the second word; and so on. Never mind how slowly you must go, even if it is only two or three words per minute. Stay behind. It will probably take quite a number of practice sessions before you can do this with any facility. After it is relatively easy, then try staying two words behind; keep this up until it is easy. Then try three words, four words, and five words. The more you practice keeping received material in mind, the easier it will be to stay behind. It will be found easier at first to copy material with which one is fairly familiar, then gradually switch to less familiar material. www.americanradiohistory.com 20 Introduction to R adio The two practice sets which are described in this chapter are of most value when you have someone with whom to practice. Automatic code machines are not recommended to anyone who can possibly obtain a companion with whom to practice, someone who is also interested in learning the code. If you are unable to enlist a code partner and have to practice Automatic Code Machines by yourself, the best way to g e t receiving practice is by the use of a tape machine (automatic code sending machine) with several practice tapes. Or you can use a set of phonograph code practice records. The records are of use only if you have a phonograph whose turntable speed is readily adjustable. The tape machine can be rented by the month for a reasonable fee. Once you can copy about 10 w.p.m. you can also get receiving practice by listening to slow sending stations on your receiver. Many amateur stations send slowly particularly when working far distant stations. When receiving conditions are particularly poor many commercial stations also send slowly, sometimes repeating every word. Until you can copy around 10 w.p.m. your receiver isn't much use, and either another operator or a machine or records are necessary for getting receiving practice after you have once memorized the code. Code Practice Sets If you don't feel too foolish doing it, you can secure a measure of code practice with the help of a partner by sending "dit-dah" messages to each other while riding to work, eating lunch, etc. It is better, however, to use a buzzer or code practice oscillator in conjunction with a regular telegraph key. As a good key may be considered an investment it is wise to make a well -made key your first purchase. Regardless of what type code practice set you use, you will need a key, and later on you will need one to key your trans- mitter. If you get a good key to begin with, you won't have to buy another one later. The key should be rugged and have fairly heavy contacts. Not only will the key stand up better, but such a key will contribute to the "heavy" type of sending so desirable for radio work. Morse (telegraph) operators use a "light" style of sending and can send somewhat faster when using this light touch. But, in radio work static and interference are often present, and a slightly heavier dot is desirable. If you use a husky key, you will find yourself automatically sending in this manner. To generate a tone simulating a code signal as heard on a receiver, either a mechanical buzzer or an audio oscillator may be used. Figure 5 shows a simple code-practice set using a buzzer which may be used directly simply by mounting the buzzer on a sounding board, or the buzzer may be used to feed from one to four pairs of conventional high -impedance phones. An example of the audio -oscillator type of code -practice set is illustrated in figures 6 and 7. An inexpensive Raytheon CK -722 transistor is used in place of the more expensive, power consuming vacuum tube. A single "pen lite" 1i-volt cell powers the unit. The coils of the earphones form the inductive portion of the resonant circuit. 'Phones having an impedance of 2000 ohms or higher should be used. Surplus type R -14 earphones also work well with this circuit. Figure 7 circuit of Figure 6 is used in this miniature transistorized code Practice oscillator. Components are mounted in a small plastic case. The transistor is The attached to a three terminal phenolic mounting strip. Sub- miniature jacks are used for the key and phones connections. A hearing aid earphone may also be used, as shown. The phone is stored in the plastic case when not in use. www.americanradiohistory.com CHAPTER TWO Direct Current Circuits so different particles, but this further subdivision can be left to quantum mechanics and atomic physics. As far as the study of electronics is concerned it is only necessary for the reader to think of the normal atom as being composed of a nucleus having a net positive charge that is exactly neutralized by the one or more orbital electrons surrounding it. The atoms of different elements differ in respect to the charge on the positive nucleus and in the number of electrons revolving around this charge. They range all the way from hydrogen, having a net charge of one on the nucleus and one orbital electron, to uranium with a net charge of 92 on the nucleus and 92 orbital electrons. The number of orbital electrons is called the atomic number of the element. All naturally occurring matter (excluding artifically produced radioactive substances) is made up of 92 fundamental constituents called elements. These elements can exist either in the free state such as iron, oxygen, carbon, copper, tungsten, and aluminum, or in chemical unions commonly called compounds. The smallest unit which still retains all the original characteristics of an element is the atom. Combinations of atoms, or subdivisions of compounds, result in another fundamental unit, the molecule. The molecule is the smallest unit of any compound. All reactive elements when in the gaseous state also exist in the molecular form, made up of two or more atoms. The nonreactive gaseous elements helium, neon, argon, krypton, xenon, and radon are the only gaseous elements that ever exist in a stable monatomic state at ordinary temperatures. From the above it must not be thought that the electrons revolve in a haphazard manner around the nucleus. Rather, the electrons in an element having a large atomic number are grouped into rings having a definite number of electrons. The only atoms in which these rings Action of the Electrons The Atom 2-1 An atom is an extremely small unit of matter there are literally billions of them making up so small a piece of material as a speck of dust. To understand the basic theory of electricity and hence of radio, we must go further and divide the atom into its main components, a positively charged nucleus and a cloud of negatively charged particles that surround the nucleus. These particles, swirling around the nucleus in elliptical orbits at an incredible rate of speed, are called orbital - are completely filled are those of the inert gases mentioned before; all other elements have one or more uncompleted rings of electrons. If the uncompleted ring is nearly empty, the element is metallic in character, being most metallic when there is only one electron in the outer ring. If the incomplete ring lacks only one or two electrons, the element is usually non- metallic. Elements with a ring about half completed will exhibit both nonmetallic and metallic characteristics; carbon, silicon, germanium, and arsenic are examples. Such elements are called semi- conductors. In metallic elements these outer ring electrons are rather loosely held. Consequently, electrons. It is upon the behavior of these electrons when freed from the atom, that depends the study of electricity and radio, as well as allied sciences. Actually it is possible to subdivide the nucleus of the atom into a dozen or 21 www.americanradiohistory.com 22 Direct Current Circuits there is a continuous helter -skelter movement of these electrons and a continual shifting from one atom to another. The electrons which move about in a substance are called free electrons, and it is the ability of these electrons to drift from atom to atom which makes possible the electric current. If the free electrons are numerous and loosely held, the element is a good conductor. On the other hand, if there are few free electrons, as is the case when the electrons in an outer ring are tightly held, the element is a poor conductor. If there are virtually no free electrons, the element is a good insulator. Conductors and Insulators 2-2 Fundamental Electrical Units and Relationships Electromotive Force: Potential Difference The free electrons in a conductor move constantly about and change their position in a haphazard manner. To produce a drift of electrons or electric current along a wire it is necessary that there be a difference in "pressure" or potential between the two ends of the wire. This potential difference can be produced by connecting a source of electrical potential to the ends of the wire. As will be explained later, there is an excess of electrons at the negative terminal of a battery and a deficiency of electrons at the positive terminal, due to chemical action. When the battery is connected to the wire, the deficient atoms at the positive terminal attract free electrons from the wire in order for the positive terminal to become neutral. The attracting of electrons continues through the wire, and finally the excess electrons at the negative terminal of the battery are attracted by the positively charged atoms at the end of the wire. Other sources of electrical potential (in addition to a battery) are: an electrical generator (dynamo), a thermocouple, an electrostatic generator (static machine), a photoelectric cell, and a crystal or piezoelectric generator. Thus it is seen that a potential difference is the result of a difference in the number of electrons between the two (or more) points in question. The force or pressure due to a potential difference is termed the electromotive force, usually abbreviated e. m. f. or E.M.F. It is expressed in units called volts. It should be noted that for there to be a potential difference between two bodies or points it is not necessary that one have a positive charge and the other a negative charge. If two bodies each have a negative THE RADIO charge, but one more negative than the other, the one with the lesser negative charge will act as though it were positively charged with respect to the other body. It is the algebraic potential difference that determines the force with which electrons are attracted or repulsed, the potential of the earth being taken as the zero reference point. The flow of electrons along a conductor due to the application of an electromotive force constitutes an electric current. This drift is in addition to the irregular movements of the electrons. However, it must not be thought that each free electron travels from one end of the circuit to the other. On the contrary, each free electron travels only a short distance before colliding with an atom; this collision generally knocking off one or more electrons from the atom, which in turn move a short distance and collide with other atoms, knocking off other electrons. Thus, in the general drift of electrons along a wire carrying an electric current, each electron travels only a short distance and the excess of electrons at one end and the deficiency at the other are balanced by the source of the e.m.f. When this source is removed the state of normalcy returns; there is still the rapid interchange of free electrons between atoms, but there is no general trend or "net movement" in either one direction or the other. The Electric Current There are two units of measure ment associated with current, and they are often confused. The rate of flou of electricity is stated in amperes. The unit of quantity is the coulomb. A coulomb is equal to 6.28 x 10" electrons, and when this quantity of electrons flows by a given point in every second, a current of one ampere is said to be flowing. An ampere is equal to one coulomb per second; a coulomb is, conversely, equal to one ampere- second. Thus we see that coulomb indicates amount, and ampere indicates rate of flow of electric current. Older textbooks speak of current flow as being from the positive terminal of the e.m.f. source through the conductor to the negative terminal. Nevertheless, it has long been an established fact that the current flow in a metallic conductor is the electronic flow from the negative terminal of the source of voltage through the conductor to the positive terminal. The only exceptions to the electronic direction of flow occur in gaseous and electrolytic conductors where the flow of positive ions toward the cathode or negative electrode constitutes a positive flow in the opposite direction to the electronic flow. (An ion is an atom, molecule, Ampere and Coulomb www.americanradiohistory.com HANDBOOK Resistance or particle which either lacks one or more electrons, or else has an excess of one or more electrons.) In radio work the terms "electron flow" and "current" are becoming accepted as being synonymous, but the older terminology is still accepted in the electrical (industrial) field. Because of the confusion this sometimes causes, it is often safer to refer to the direction of electron flow rather than to the direction of the "current." Since electron flow consists actually of a passage of negative charges, current flow and algebraic electron flow do pass in the same direction. The flow of current in a material depends upon the ease with which electrons can be detached from the atoms of the material and upon its molecular structure. In other words, the easier it is to detach electrons from the atoms the more free electrons there will be to contribute to the flow of current, and the fewer collisions that occur between free electrons and atoms the greater will be the total electron flow. The opposition to a steady electron flow is called the resistance of a material, and is one of its physical properties. The unit of resistance is the ohm. Every 23 TABLE OF RESISTIVITY ' Material Aluminum Bross Cadmium Chromium Copper Iron Silver Zinc Nichrome Const Manganin Monet esist vny in Ohms per Temp. Coeff. of resistance per =C at 20° C. Circular Mil -Foot 0.0049 0.003 to 0.007 0.0038 0.00 0.0039 0.006 0.004 0.0035 0.0002 17 45 46 16 10.4 59 9.8 36 650 0.00001 0.00001 0.0019 295 290 255 FIGURE 1 Resistance substance has a specific resistance, usually expressed as ohms per mil -foot, which is determined by the material' s molecular structure and temperature. A mil -foot is a piece of material one circular mil in area and one foot long. Another measure of resistivity frequently used is expressed in the units microhms per centimeter cube. The resistance of a uniform length of a given substance is directly proportional to its length and specific resistance, and inversely proportional to its cross- sectional area. A wire with a certain resistance for a given length will have twice as much resistance if the length of the wire is doubled. For a given length, doubling the cross -sectional area of the wire will halve the resistance, while doubling the diameter will reduce the resistance to one fourth. This is true since the cross -sectional area of a wire varies as the square of the diameter. The relationship between the resistance and the linear dimensions of a conductor may be expressed by the following equation: R R = r = l= A = Conductors and resistance in ohms resistivity in Ohms per mil-foot length of conductor in feet cross - sectional area in circular mils In the molecular structure of glass, porcelain, and mica all electrons are tightly held within their orbits and there are comparatively few free electrons. This type of substance will conduct an electric current only with great difficulty and is known as an insulator. An insulator is said to have a high electrical resistance. On the other hand, materials that have a large number of free electrons are known as conductors. Alost metals, those elements which have only one or two electrons in their outer ring, are good conductors. Silver, copper, and aluminum, in that order, are the best of the common metals used as conductors and are said to have the greatest conductivity, or lowest resistance to the flow of an electric current. These units are the volt, Fundamental Insulators many materials such as the ampere, and the ohm. They were mentioned in the preceding paragraphs, but were not completely defined in terms of fixed, known quantities. The fundamental unit of current, or rate of flow of electricity is the ampere. A current of one ampere will deposit silver from a specified solution of silver nitrate at a rate of 1.118 milligrams per second. Electrical Units =-rl A Where The resistance also depends upon temperature, increasing with increases in temperature for most substances (including most metals), due to increased electron acceleration and hence a greater number of impacts between electrons and atoms. However, in the case of some substances such as carbon and glass the temperature coefficient is negative and the resistance decreases as the temperature increases. This is also true of electrolytes. The temperature may be raised by the external application of heat, or by the flow of the current itself. In the latter case, the temperature is raised by the heat generated when the electrons and atoms collide. www.americanradiohistory.com 24 Direct Current Circuits THE RADIO 1111111111111111 lu 1 Figure 2 TYPICAL RESISTORS Shown above are various types of resistors used in electronic circuits. The larger units are power resistors. On the left is a variable power resistor. Three precision -type resistors ore shown in the tenter with two small composition resistors beneath them. At the right is o composition -type potentiometer, used for audio circuitry. The international standard for the ohm is the resistance offered by a uniform column of mercury at 0°C., 14.4521 grams in mass, of constant cross - sectional area and 106.300 centimeters in length. The expression megohm (1,000,000 ohms) is also sometimes used when speaking of very large values of resistance. A volt is the e.m.f. that will produce a current of one ampere through a resistance of one ohm. The standard of electromotive force is the Weston cell which at 20 °C. has a potential of 1.0183 volts across its terminals. This cell is used only for reference purposes in a bridge circuit, since only an infinitesimal -vw-v- RESISTANCE Ri vor CONDUCTORS B2 _- BATTERY E Figure 3 SIMPLE SERIES CIRCUITS At (A) the battery is in series with a single resistor. At (B) the battery is in series with two resistors, the resistors themselves being in series. The arrows indicate the direction of electron flow. amount of current may be drawn from it without disturbing its characteristics. The relationship between the electromotive force (voltage), the flow of current (amperes), and the resistance which impedes the flow of current (ohms), is very clearly expressed in a simple but highly valuable law known as Ohm's laun. This law states that the current in amperes is equal to the voltage in volts divided by the resistance in ohms. Expressed as an equation: Ohm's Law I =-RE If the voltage (E) and resistance (R) are known, the current (I) can be readily found. If the voltage and current are known, and the resistance is unknown, the resistance (R) is E equal to . When the voltage is the un- known quantity, it can be found by multiplying I x R. These three equations are all secured from the original by simple transposition. The expressions are here repeated for quick reference: E I =R www.americanradiohistory.com R=-E I E = IR Resistive Circuits HANDBOOK Figure 4 SIMPLE PARALLEL CIRCUIT The two resistors RI and R2 are said to be in parallel since the flow of current is offered two parallel paths. An electron leaving point A will pass either through R1 or R2, but not through both, to reach the positive terminal of the battery. If a large number of lectrons are considered, the greater number will pass through whichever of the two resistors has the lower resistance. where I is the current in amperes, R is the resistance in ohms, E is the electromotive force in volts. Application of All electrical circuits fall in- Ohm's Law to one of three classes: series circuits, parallel circuits, and series -parallel circuits. A series circuit is one in which the current flows in a single continuous path and is of the same value at every point in the circuit (figure 3). In a parallel circuit there are two or more current paths between two points in the circuit, as shown in figure 4. Here the current divides at A, part going through R, and part through R2i and combines at B to return to the battery. Figure 5 shows a series -parallel circuit. There are two paths between points A and B as in the parallel circuit, and in addition there are two resistances in series in each branch of the parallel combination. Two other examples of series -parallel arrangements appear in figure 6. The way in which the current splits to flow through the parallel branches is shown by the arrows. In every circuit, each of the parts has some resistance: the batteries or generator, the connecting conductors, and the apparatus itself. Thus, if each part has some resistance, no matter how little, and a current is flowing through it, there will be a voltage drop across it. In other words, there will be a potential difference between the two ends of the circuit element in question. This drop in voltage is equal to the product of the current and the resistance, hence it is called the IR drop. The source of voltage has an internal resistance, and when connected into a circuit so that current flows, there will be an IR drop in the source just as in every other part of the circuit. Thus, if the terminal voltage of the source could be measured in a way that would cause no current to flow, it would be found to be more than the voltage measured when a current flows by the amount of the IR drop 25 Figure 5 SERIES-PARALLEL CIRCUIT In this type of circuit the resistors are arranged in series groups, and these serlesed groups ore then placed in parallel. in the source. The voltage measured with no current flowing is termed the no load voltage; that measured with current flowing is the load voltage. It is apparent that a voltage source having a low internal resistance is most de- sirable. The current flowing in a series circuit is equal to the voltage impressed divided by the total resistance across which the voltage is impressed. Since the same current flows through every part of the circuit, it is merely necessary to add all the individual resistances to obtain the total resistance. Expressed as a formula: Resistances in Series +... +RN . Riotai =RI +R2 +R, if the resistances happened to be all the same value, the total resistance would be the resistance of one multiplied by the number of resistors in the circuit. Of course, Consider two resistors, one of 100 ohms and one of 10 ohms, connected in parallel as in figure 4, with a voltage of 10 volts applied across each resistor, so the current through each can be easily calculated. Resistances in Parallel E I= -R E = 10 volts I, = R = 100 ohms E = 10 volts R ohms 10 Total current 10 = 0.1 ampere 100 -= 1.0 ampere 10 I2 = 10 = I, + 12 = 1.1 ampere Until it divides at A, the entire current of 1.1 amperes is flowing through the conductor from the battery to A, and again from B through the conductor to the battery. Since this is more current than flows through the smaller resistor it is evident that the resistance of the parallel combination must be less than 10 ohms, the resistance of the smaller resistor. We can find this value by applying Ohm's law. www.americanradiohistory.com Direct Current Circuits 26 THE R=-E RADIO A I E = 10 I = 1.1 volts amperes 10 R = 1.1 9.09 ohms The resistance of the parallel combination is 9.09 ohms. Mathematically, we can derive a simple formula for finding the effective resistance of two resistors connected in parallel. This formula is: - R where R R, R2 R, x R, +R, is the unknown resistance, is the resistance of the first resistor, is the resistance of the second resistor. If the effective value required is known, and it is desired to connect one unknown resistor in parallel with one of known value, the following transposition of the above formula will simplify the problem of obtaining the unknown value: R2 where R, x R - R, -R is the effective value required, R, is the known resistor, R2 is the value of the unknown resistance necessary to give R when in parallel with R,. R The resultant value of placing a number of unlike resistors in parallel is equal to the reciprocal of the sum of the reciprocals of the various resistors. This can be expressed as: R= 1 - +1 R, 1 R, Figure 6 OTHER COMMON SERIES -PARALLEL CIRCUITS R, -+... R, 1 + - resistors connected in parallel is always less than the value of the lowest resistance in more the combination. It is well to bear this simple rule in mind, as it will assist greatly in approximating the value of paralleled resistors. To find the total resistance of several resistors connected in series -parallel, it is usually easiest to apply either the formula for series resistors or the parallel resistor formula first, in order to reduce the original arrangement to a simpler one. For instance, in figure 5 the series resistors should be added in each branch, then there will be but two resistors in parallel to be calculated. Similarly in figure 7, although here there will be three parallel resistors after adding the series resistors in each branch. In figure 6B the paralleled resistors should be reduced to the equivalent series value, and then the series resistance values can be added. Resistances in series -parallel can be solved by combining the series and parallel formulas into one similar to the following (refer to figure 7): Resistors in Series Parallel 1 R. R1 The effective value of placing any number of unlike resistors in parallel can be determined from the above formula. However, it is commonly used only when there are three or more resistors under consideration, since the simplified formula given before is more convenient when only two resistors are being used. From the above, it also follows that when two or more resistors of the same value are placed in parallel, the effective resistance of the paralleled resistors is equal to the value of one of the resistors divided by the number of resistors in parallel. The effective value of resistance of two or R,+ +-- 1 1 R, R, + R, 1 Rs +R6+R, A voltage divider is exactly what its name implies: a resistor or a series of resistors connected across a source of voltage from which various lesser values of voltage may be obtained by connection to various points along the resistor. A voltage divider serves a most useful purpose in a radio receiver, transmitter or amplifier, because it offers a simple means of obtaining plate, screen, and bias voltages of different values from a common power supply Voltage Dividers www.americanradiohistory.com HANDBOOK Voltage Divider 27 BLEEDER CURRENT i__ FLOWS BETWEEN POINTS A AND B EATERNAL LOAD Figure 7 ANOTHER TYPE OF SERIES -PARALLEL CIRCUIT Figure 8 SIMPLE VOLTAGE DIVIDER source. It may also be used to obtain very low voltages of the order of .01 to .001 volt with a high degree of accuracy, even though a means of measuring such voltages is lacking. The procedure for making these measurements can best be given in the following example. Assume that an accurately calibrated voltmeter reading from 0 to 150 volts is available, and that the source of voltage is exactly 100 volts. This 100 volts is then impressed through a resistance of exactly 1,000 ohms. It will, then, be found that the voltage along various points on the resistor, with respect to the grounded end, is exactly proportional to the resistance at that point. From Ohm's law, the current would be 0.1 ampere; this current remains unchanged since the original value of resistance (1,000 ohms) and the voltage source (100 volts) are unchanged. Thus, at a 500 ohm point on the resistor (half its entire resistance), the voltage will likewise be halved or reduced to 50 volts. The equation (E = I x R) gives the proof: E = 500 x 0.1 = 50. At the point of 250 ohms on the resistor, the voltage will be one -fourth the total value, or 25 volts (E = 250 x 0.1 = 25). Continuing with this process, a point can be found where the resistance measures exactly 1 ohm and where the voltage equals 0.1 volt. It is, therefore, obvious that if the original source of voltage and the resistance can be measured, it is a simple matter to predetermine the voltage at any point along the resistor, provided that the current remains constant, and provided that no current is taken from the tap -on point unless this current is taken into consideration. Proper design of a voltage divider for any type of radio equipment is a relatively simple matter. The first consideration is the amount of "bleeder current" to be drawn. In addition, it is also necessary that the desired voltage and the exact current at each tap on the voltage divider be known. Figure 8 illustrates the flow of current in a simple voltage divider and load circuit. The light arrows indicate the flow of bleeder current, while the heavy arrows indicate the flow of the load current. The design of a combined Voltage Divider Calculations CIRCUIT indicate the manner in which the current flow divides between the voltage divider itslf and th externo! load circuit. The arrows bleeder resistor and voltage divider, such as is commonly used in radio equipment, is illustrated in the following example: A power supply delivers 300 volts and is conservatively rated to supply all needed current for the receiver and still allow a bleeder current of 10 milliamperes. The following voltages are wanted: 75 volts at 2 milliamperes for the detector tube, 100 volts at 5 milliamperes for the screens of the tubes, and 250 volts at 20 milliamperes for the plates of the tubes. The required voltage drop across R, is 75 volts, across R, 25 volts, across R, 150 volts, and across R, it is 50 volts. These values are shown in the diagram of figure 9. The respective current values are also indicated. Apply Ohm's law: E R, 75 = = 7,500 ohms. 01 R, R, = R, E 25 I 012 E -_ I -= 150 2,083 ohms. 8,823 ohms. .017 50 =-E = .037 = 1,351 ohms. RTotal = 7,500 + 2,083 + 8,823 + 1,351 = 19,757 ohms. A 20,000 -ohm resistor with three sliding taps will be of the approximately correct size, and would ordinarily be used because of the diffi- culty in securing four separate resistors of the exact odd values indicated, and because no adjustment would be possible to compensate for any slight error in estimating the probable currents through the various taps. When the sliders on the resistor once are set to the proper point, as in the above ex- www.americanradiohistory.com Direct Current Circuits 28 THE -2 10 + 2 +5 +20 AMPS 1M RI MA. 50 VOLTS DROP 0V 0MA A -2-AMPS L1, R2 10 +2 +5 MA 150 VOLTS DROP AMPi - 1 300 VOLTS +2 -1 (-/ 10 RADIO MA. 25 VOLTS DROP Figure 10 ILLUSTRATING KIRCHHOFF'S l R CURRENT10 .A.J BLEEDE75 VOLTS D,ROP POWER SUPPLY Figure - LOA D - - - - - 9 MORE COMPLEX VOLTAGE DIVIDER The method for computing the values of the resistors is discussed in the accompanying text. FIRST LAW The current flowing toward point "A" is to the current flowing away from point qual "A." - sum of all currents flowing toward and away from the point taking signs into account is equal to zero. Such a sum is known as an algebraic sum; such that the law can be stated thus: The algebraic sum of all tive, the - ample, the voltages will remain constant at the values shown as long as the current remains a constant value. currents entering and leaving a point is zero. Figure 10 illustrates this first law. Since the effective resistance of the network of resistors is 5 ohms, it can be seen that 4 amperes flow One of the serious disadvanrages of the voltage divider becomes evident when the the current drawn fromone of the taps changes. It is obvious that the voltage drops are interdependent and, in turn, the individual drops are in proportion to the current which flows through the respective sections of the divider resistor. The only remedy lies in providing a heavy steady bleeder current in order to make the individual currents so small a part of the total current that any change in current will toward point A, and 2 amperes flow away through the two 5 -ohm resistors in series. The remaining 2 amperes flow away through the 10ohm resistor. Thus, there are 4 amperes flowing to point A and 4 amperes flowing away from the point. If R is the effective resistance of the network (5 ohms), R, = 10 ohms, R, = 5 ohms, R, = 5 ohms, and E = 20 volts, we can set up the following equation: Disadvantages of Voltage Dividers result in only a slightchange in voltage. This can seldom be realized in practice because of the excessive values of bleeder current which would be required. Kirchhoff's Laws Ohm's law is all that is necessary to calculate the values in simple circuits, such as the preceding examples; but in more complex problems, involving several loops or more than one voltage in the same closed circuit, the use of Kirchhoff's laws will greatly simplify the calculations. These laws are merely rules for applying Ohm's law. Kirchhoff's first law is concerned with net current to a point in a circuit and states that: At any point in a circuit the current flowing toward the point is equal to the current flowing away from the point. Stated in another way: if currents flowing to the point are considered positive, and those flowing from the point are considered nega- E E R R, E 20 20 5 10 R2 +R, =0 20 5 +5 4 -2 -2 =0 Kirchhoff's second law is concerned with net voltage drop around a closed loop in a circuit and states that: In any closed path or loop in a circuit the sum of the IR drops must equal the sum of the applied e.m. f.'s. The second law also may be conveniently stated in terms of an algebraic sum as: The algebraic sum of all voltage drops around a closed path or loop in a circuit is zero. The applied e.m.f.'s (voltages) are considered positive, while IR drops taken in the direction of current flow (including the internal drop of the sources of voltage) are considered negative. Figure 11 shows an example of the applica- tion of Kirchhoff's laws to a comparatively simple circuit consisting of three resistors and www.americanradiohistory.com Kirchoff's Laws HANDBOOK 29 1 volt forces a current of 1 ampere through a circuit. The power in a resistive circuit is equal to the product of the voltage applied across, and the current flowing in, a given circuit. Hence: P (watts) = E (volts) x I (amperes). Since it is often convenient to express power in terms of the resistance of the circuit and the current flowing through it, a substitution of IR for E (E = IR) in the above formula gives: P = IR x I or P = 12R. In terms of voltage and resistance, P = E' /R. Here, I = E/R and when this is substituted for I the original formula becomes P = E x E /R, or P = E' /R. To repeat these three expressions: an e.m.f. of 1. 2. SET VOLTAGE DROPS AROUND EACH LOOP EQUAL TO ZERO. 1121DHMS)+2(t -12)+3 =0 -6+2 (12-11) +312 °0 (FIRST LOOP) (SECOND LOOP) SIMPLIFY 211+211-212+3.0 411 +3 - 2 1 21a- 2It+31z -6 =0 512- 211 -6 =0 2 211+6 5 3. 411 +3 2 4 5 - I z P = EQUATE 2It +6 - 5 SIMPLIFY and 2011+15= 411 +12 11 ß-t6 AMPERE I RE- SUBSTITUTE Iz- 23 2 EI, P = I2R, and P = E2 /R, where P is the power in watts, E is the electromotive force in volts, t 2 I á AMPERE Figure 11 ILLUSTRATING KIRCHHOFF'S SECOND LAW The voltage drop around any closed loop In a network Is qual to zero. two batteries. First assume an arbitrary direction of current flow in each closed loop of the circuit, drawing an arrow to indicate the assumed direction of current flow. Then equate the sum of all IR drops plus battery drops around each loop to zero. You will need one equation for each unknown to be determined. Then solve the equations for the unknown currents in the general manner indicated in figure 11. If the answer comes out positive the direction of current flow you originally assumed was correct. If the answer comes out negative, the current flow is in the opposite direction to the arrow which was drawn originally. This is illustrated in the example of figure 11 where the direction of flow of I, is opposite to the direction assumed in the sketch. In order to cause electrons Resistive Circuits to flow through a conductor, constituting a current flow, it is necessary to apply an electromotive force (voltage) across the circuit. Less power is expended in creating a small current flow through a given resistance than in creating a large one; so it is necessary to have a unit of power as a reference. The unit of electrical power is the watt, which is the rate of energy consumption when is the current in amperes. To apply the above equations to a typical problem: The voltage drop across a cathode resistor in a power amplifier stage is 50 volts; the plate current flowing through the resistor is 150 milliamperes. The number of watts the resistor will be required to dissipate is found from the formula: P = El, or 50 x .150 = 7.5 watts (.150 amperes is equal to 150 milliamperes). From the foregoing it is seen that a 7.5 -watt resistor will safely carry the required current, yet a 10- or 20 -watt resistor would ordinarily be used to provide a safety factor. In another problem, the conditions being similar to those above, but with the resistance (R = 333`/2 ohms), and current being the known factors, the solution is obtained as follows: P = I2R = .0225 x 333.33 = 7.5. If only the voltage and resistance are known, P = E2 /R = 2500/333.33 = 7.5 watts. It is seen that all three equations give the same results; the selection of the particular equation depends only upon the known factors. It is important to remember that power (expressed in watts, horsepower, etc.), represents the rate of energy consumption or the rate of doing work. But when we pay our electric bill Power, Energy and Work Power in Figure 12 MATCHING OF RESISTANCES RL I To deliver the greatest amount of power to the load, the load resistance RL should be equal to the Internal reslstonce of the battery RI. www.americanradiohistory.com 30 Direct Current Circuits THE RADIO is said to have a certain capacitance. The energy stored in an electrostatic field is expressed in joules (watt seconds) and is equal to CE' /2, where C is the capacitance in farads (a unit of capacitance to be discussed) and E is the potential in volts. The charge is equal to CE, the charge being expressed in coulombs. metallic plates separated from each other by a thin layer of insulating material (called a dielectric, in this case), becomes a capacitor. When a source of d-c potential is momentarily applied across these plates, they may be said to become charged. If the same two plates are then joined together momentarily by means of a switch, the capacitor will discharge. When the potential was first applied, electrons immediately flowed from one plate to the other through the battery or such source of d -c potential as was applied to the capacitor plates. However, the circuit from plate to plate in the capacitor was incomplete (the two plates being separated by an insulator) and thus the electron flow ceased, meanwhile establishing a shortage of electrons on one plate and a surplus of electrons on the other. Remember that when a deficiency of electrons exists at one end of a conductor, there is always a tendency for the electrons to move about in such a manner as to re- establish a state of balance. In the case of the capacitor herein discussed, the surplus quantity of electrons on one of the capacitor plates cannot move to the other plate because the circuit has been broken; that is, the battery or d -c potential was removed. This leaves the capacitor in a charged condition; the capacitor plate with the electron deficiency is positively charged, the other plate being negative. In this condition, a considerable stress exists in the insulating material (dielectric) which separates the two capacitor plates, due to the mutual attraction of two unlike potentials on the plates. This stress is known as electrostatic energy, as contrasted with electromagnetic energy in the case of an inductor. This charge can also be called potential energy because it is capable of performing work when the charge is released through an external circuit. The charge is proportional to the voltage but the energy is proportional to the voltage squared, as shown in the following Capacitance and Capacitors sm. nErg Figure 13 TYPICAL CAPACITORS large units ore high value filter capaci- The two tors. Shown beneath these ore various types of by -pass capacitors for r-f and audio application. to the power company we have purchased a specific amount of energy or work expressed in the common units of kilowatt- hours. Thus rate of energy consumption (watts or kilowatts) multiplied by time (seconds, minutes or hours) gives us total energy or work. Other units of energy are the watt- second, BTU, calorie, erg, and joule. Heating Effect Heat is generated when a source of voltage causes a current to flow through a resistor (or, for that matter, through any conductor). As explained earlier, this is due to the fact that heat is given off when free electrons collide with the atoms of the material. More heat is generated in high resistance materials than in those of low resistance, since the free electrons must strike the atoms harder to knock off other electrons. As the heating effect is a function of the current flowing and the resistance of the circuit, the power expended in heat is given by the second formula: P = I'R. 2 -3 Electrostatics - Capacitors Electrical energy can be stored in an electrostatic field. A device capable of storing energy in such a field is called capacitor (in earlier usage the term condenser was frequently used but the IRE standards call for the use of capacitor instead of condenser) and Two analogy. The charge represents a definite amount of electricity, or a given number of electrons. The potential energy possessed by these electrons depends not only upon their number, but also upon their potential or voltage. Compare the electrons to water, and two capacitors to standpipes, a www.americanradiohistory.com 1 fifd. capacitor to Capacitance HANDBOOK ROSTATIC A- EILECT ELD - SHORTAGE OF ELECTRONS 1 SURPLUS OF ELECTRONS 1 Figure 14 SIMPLE CAPACITOR Illustrating the imaginary lines of force repre Renting the paths along which the repelling force If the external circuit of the two capacitor plates is completed by joining the terminals together with a piece of wire, the electrons will rush immediately from one plate to the other through the external circuit and establish a state of equilibrium. This latter phenomenon explains the discharge of a capacitor. The amount of stored energy in a charged capacitor is dependent upon the charging potential, as well as a factor which takes into account the size of the plates, dielectric thickness, nature of the dielectric, and the number of plates. This factor, which is determined by the foregoing, is called the capacitanceof a capacitor and is expressed in farads. The farad is such a large unit of capacitance that it is rarely used in radio calculations, and the following more practical units The Unit of Capacitance: The Farad have, therefore, been chosen. farad, - CxE' 2 x 1,000,000 This storage of energy in a capacitor is one of its very important properties, particularly in those capacitors which are used in power supply filter circuits. a standpipe having a cross section of 1 square inch and a 2 pfd. capacitor to a standpipe having a cross section of 2 square inches. The charge will represent a given volume of water, as the "charge" simply indicates a certain number of electrons. Suppose the water is equal to 5 gallons. Now the potential energy, or capacity for doing work, of the 5 gallons of water will be twice as great when confined to the 1 sq. in. standpipe as when confined to the 2 sq. in. standpipe. Yet the volume of water, or "charge" is the same in either case. Likewise a 1 pfd. capacitor charged to 1000 volts possesses twice as much potential energy as does a 2 pfd. capacitor charged to 500 volts, though the charge (expressed in coulombs: Q = CE) is the same in either case. a micro-microlarad = one - millionth of one millionth of a farad, or 10'E' farads. Stored energy in joules of the electrons would act on o free electron located between the two capacitor plates. micro farad = 1 /1,000,000 of .000001 farad, or 10-6 farads. micro- microfarad = 1 /1,000,000 of a micro farad, or .000001 microfarad, or 10'6 micro farads. If the capacitance is to be expressed in microfarads in the equation given for energy storage, the factor C would then have to be divided by 1,000,000, thus: CHARGING CURRENT 1 31 or Although any substance which has the characteristics of a good insulator may be used as a dielectric material, commercially manufactured capacitors make use of dielectric materials which have been selected because their characteristics are particularly suited to the job at hand. Air is a very good dielectric material, but an air - spaced capacitor does not have a high capacitance since the dielectric constant of air is only slightly greater than one. A group of other commonly used dielectric mate ials is listed in figure 15. Certain materials, such as bakelite, lucite, and other plastics dissipate considerable energy when used as capacitor dielectrics. Dielectric Materials 1 DIELECTRIC CONSTANT MATERIAL 1O ANILINE- FORMALDEHYDE RESIN BARIUM TITANATE MC. POWER FACTOR 1O MC. 3 4 0.004 1200 1.0 .67 CASTOR OIL 3.7 CELLULOSE ACETATE GLASS.WINDOW 6 GLASS, PYREX FLUOROTHENE XEL -F METHYL - METHACRYLATE LUCITE MICA MYCALEX, MYKROY PHENOL -FORMALDEHYDE, LOW-LOSS YELLOW PHENOL -FORMALDEHYDE BLACK BAKELITE PORCELAIN POLYETHYLENE POLYSTYRENE QUARTZ FUSED RUBBER, HARD-EBONITE STEATITE SULFUR TEFLON TITANIUM DIOXIDE TRANSFORMER OIL UREA -FORMALDEHYDE VINYL RESINS WOOD. MAPLE - -6 0.04 POOR SOFTENING POINT FAHRENHEIT 260 - IRO 2000 4.5 0.02 U.S 0.6 - 2.6 5.4 7.0 5.0 0.007 160 5.5 0.03 7.0 25 0.005 0.0003 0.0002 0 0002 0.007 0.003 0.003 0.02 0.0006 0.003 0.05 0.02 2 2.55 4.2 2.6 6.1 3.6 2.1 100 -175 2.2 5.0 4.0 . FIGURE 15 0.0003 0.002 0.015 POOR 650 270 330 _2600 220 175' 2600 150 2700' 236 - 2700 260 200 Direct Current Circuits 34 1 C 1 1 1 C, 1 1 1 1 C, C, C, C, capacitor is connected into a direct -current circuit, it will block the d.c., or stop the flow of current. Beyond the initial movement of electrons during the period when the capacitor is being charged, there will be no flow of current because the circuit is effectively broken by the dielectric of the capacitor. Strictly speaking, a very small current may actually flow because the dielectric of the capacitor may not be a perfect insulator. This minute current flow is the leakage current previously referred to and is dependent upon the internal d -c resistance of the capacitor. This leakage current is usually quite noticeable in most types of electrolytic capacitors. When an alternating current is applied to a capacitor, the capacitor will charge and discharge a certain number of times per second in accordance with the frequency of the alternating voltage. The electron flow in the charge and discharge of a capacitor when an a-c potential is applied constitutes an alternating current, in effect. It is for this reason that a capacitor will pass an alternating current yet offer practically infinite opposition to a direct current. These two properties are repeatedly in evidence in a radio circuit. Capacitors in and D -C EQUAL RESISTANCE EQUAL CAPACITANCE When a A -C Circuits good paper dielectric filter capacitor has such a high internal resistance (inin Series dicating a good dielectric) that the exact resistance will vary considerably from capacitor to capacitor even though they are made by the same manufacturer and are of the same rating. Thus, when 1000 volts d.c. is connected across two 1-pfd. 500-volt capacitors in series, the chances are that the voltage will divide unevenly and one capacitor will receive more than 500 volts and the other less than 500 volts. Voltage Rating of Capacitors RADIO 1 - +- - +- - +C, THE Any connecting a half 1 -watt carbon resistor across each capacitor, the voltage will be equalized because the resistors act as a voltage divider, and the internal resistances of the capacitors are so much higher (many megohms) that they have but little effect in disturbing the voltage divider balance. Carbon resistors of the inexpensive type are not particularly accurate (not being designed for precision service); therefore it is Voltage Equalizing Resistors By megohm Figure 18 SHOWING THE USE OF VOLTAGE EQUALIZING RESISTORS ACROSS CAPACITORS CONNECTED IN SERIES advisable to check several ohmmeter to find two that possible in resistance. The is unimportant, just so it is two resistors used. on an accurate are as close as exact resistance the same for the capacitors are connected in series, alternating voltage pays no heed to the relatively high internal resistance of each capacitor, but divides across the capacitors in inverse proportion to the capacitance. Because, in addition to the d.c. across a capacitor in a filter or audio amplifier circuit there is usually an a -c or a -f voltage component, it is inadvisable to series -connect capacitors of unequal capacitance even if dividers are provided to keep the d.c. within the ratings of the individual capacitors. For instance, if a 500 -volt 1 -µfd. capacitor is used in series with a 4-pfd. 500 -volt capacitor across a 250 -volt a -c supply, the 1 -µfd. capacitor will have 200 volts a.c. across it and the 4-pfd. capacitor only 50 volts. An equalizing divider to do any good in this case would have to be of very low resistance because of the comparatively low impedance of the capacitors to a.c. Such a divider would draw excessive current and be impracticable. The safest rule to follow is to use only capacitors of the same capacitance and voltage rating and to install matched high resistance proportioning resistors across the various capacitors to equalize the d-c voltage drop across each capacitor. This holds regardless of how many capacitors are series -connected. Capacitors in Series on A.C. When two Electrolytic capacitors use a very thin film of oxide as the dielectric, and are polarized; that is, they have a positive and a negative terminal which must be properly connected in a circuit; otherwise, the oxide will break down and the capacitor will overheat. The unit then will no longer be of service. When electrolytic capacitors are connected in series, the positive terminal is always connected to the positive lead of the power supply; the negative terminal of Electrolytic Capacitors HANDBOOK M the capacitor connects to the positive terminal of the next capacitor in the series combination. The method of connection for electrolytic capacitors in series is shown in figure 18. Electrolytic capacitors have very low cost per microfarad of capacity, but also have a large power factor and high leakage; both dependent upon applied voltage, temperature and the age of the capacitor. The modern electrolytic capacitor uses a dry paste electrolyte embedded in a gauze or paper dielectric. Aluminium foil and the dielectric are wrapped in a circular bundle and are mounted in a cardboard or metal box. Etched electrodes may be employed to increase the effective anode area, and the total capacity of the unit. The capacity of an electrolytic capacitor is affected by the applied voltage, the usage of the capacitor, and the temperature and humidity of the environment. The capacity usually drops with the aging of the unit. The leakage current and power factor increase with age. At high frequencies the power factor becomes so poor that the electrolytic capacitor acts as a series resistance rather than as a capacity. Magnetism 2 -4 and Electromagnetism The common bar or horseshoe magnet is familiar to most people. The magnetic field which surrounds it causes the magnet to attract other magnetic materials, such as iron nails or tacks. Exactly the same kind of magnetic field is set up around any conductor carrying a current, but the field exists only while the current is flowing. Magnetic Fields Before a potential, or voltage, is applied to a con- ductor there is no external field, because there is no general movement of the electrons in one direction. However, the electrons do progressively move along the conductor when an e.m.f. is applied, the direction of motion depending upon the polarity of the e.m.f. Since each electron has an electric field about it, the flow of electrons causes these fields to build up into a resultant external field which acts in a plane at right angles to the direction in which the current is flowing. This field is known as the magnetic field. The magnetic field around a current-carrying conductor is illustrated in figure 19. The direction of this magnetic field depends entirely upon the direction of electron drift or current flow in the conductor. When the flow is toward the observer, the field about the conductor is clockwise; when the flow is away from the observer, the field is counter- clockwise. This is easily remembered if the left hand is clenched, with the thumb outstretched agnetism 35 ELECTRON DRIFT .."-SWITCH Figure 19 LEFT -HAND RULE Showing the direction of the magnetic lines of force produced around a conductor carrying an electric current. and pointing in the direction of electron flow. The fingers then indicate the direction of the magnetic field around the conductor. Each electron adds its field to the total external magnetic field, so that the greater the number of electrons moving along the conductor, the stronger will be the resulting field. One of the fundamental laws of magnetism is that like poles repel one another and unlike poles attract one another. This is true of current- carrying conductors as well as of permanent magnets. Thus, if two conductors are placed side by side and the current in each is flowing in the same direction, the magnetic fields will also be in the same direction and will combine to form a larger and stronger field. If the current flow in adjacent conductors is in opposite directions, the magnetic fields oppose each other and tend to cancel. The magnetic field around a conductor may be considerably increased in strength by winding the wire into a coil. The field around each wire then combines with those of the adjacent turns to form a total field through the coil which is concentrated along the axis of the coil and behaves externally in a way similar to the field of a bar magnet. If the left hand is held so that the thumb is outstretched and parallel to the axis of a coil, with the fingers curled to indicate the direction of electron flow around the turns of the coil, the thumb then points in the direction of the north pole of the magnetic field. The Magnetic In the magnetic circuit, the units which correspond to current, voltage, and resistance in the electrical circuit are flux, magneto motive force, and reluctance. Circuit Flux, Flux Density is made up of a drift of electrons, so is a magnetic field made up of lines of force, and the total number of lines of force in a given magnetic circuit is termed the flux. The flux depends upon the material, cross section, and length of the magnetic circuit, and it varies directly as the current flowing in the circuit. As a current www.americanradiohistory.com Direct Current Circuits 36 The unit of flux is the maxwell, and the symbol is the Greek letter cp (phi). Flux density is the number of lines of force per unit area. It is expressed in gauss if the unit of area is the square centimeter (1 gauss = 1 line of force per square centimeter), or in lines per square inch. The symbol for flux density is B if it is expressed in gausses, or B if expressed in lines per square Inch. The force which produces a flux in a magnetic circuit is called magnetomotive force. It is abbreviated m.m.f. and is designated by the letter F. The unit of magnetomotive force is the gilbert, which is equivalent to 1.26 x NI, where N is the number of turns and I is the current flowing in the circuit in amperes. The m.m.f. necessary to produce a given flux density is stated in gilberts per centimeter (oersteds) (H), or in ampere -turns per inch (H). magnetomotive Force Magnetic reluctance corresponds to electrical resistance, and is the property of a material that opposes the creation of a magnetic flux in the material. It is expressed in rels, and the symbol is the letter R. A material has a reluctance of 1 rel when an m.m.f. of 1 ampere -turn (NI) generates a flux of 1 line of force in it. Combinations of reluctances are treated the same as resistances in finding the total effective reluctance. The specific reluctance of any substance is its reluctance per unit volume. Except for iron and its alloys, most common materials have a specific reluctance very nearly the same as that of a vacuum, which, for all practical purposes, may be considered the same as the specific reluctance of air. Reluctance Ohm's Law for The relations between flux, Magnetic Circuits magnetomotive force, and reluctance are exactly the same as the relations between current, voltage, and resistance in the electrical circuit. These can be stated as follows: F F R THE duce in air. It may be expressed by the ratio B/H or B/H. In other words, B ç = flux, F = B or R H H where p is the premeability, B is the flux density in gausses, B is the flux density in lines per square inch, H is the m.m.f. in gilberts per centimeter (oersteds), and H is the m.m.f. in ampere -turns per inch. These relations may also be stated as follows: B H=- or fi B H=-, f and B=Hit or B= Permeability is similar to electric conductivity. There is, however, one important difference: the permeability of magnetic materials is not independent of the magnetic current (flux) flowing through it, although electrical conductivity is substantially independent of the electric current in a wire. When the flux density of a magnetic conductor has been increased to the saturation point, a further increase in the magnetizing force will not produce a corresponding increase in flux density. Saturation magnetic circuit a magnetization curve may be drawn for a given unit of material. Such a curve is termed a B -H curve, and may be determined by experiment. When the current in an iron core coil is first applied, the relation between the winding current and the core flux is shown at A -B in figure 20. If the current is then reduced to zero, reversed, brought back again to zero and reversed to the Calculations To simplify calculations, F=chR - MAGNETIZING FORCE m.m.f., and R = H reluctance. Permeability expresses the ease with which a magnetic field may be set up in a material as compared with the effort required in the case of air. Iron, for example, has a permeability of around 2000 times that of air, which means that a given amount of magnetizing effect produced in an iron core by a current flowing through a coil of wire will produce 2000 times the flux density that the same magnetizing effect would pro- Hµ It can be seen from the foregoing that permeability is inversely proportional to the specific reluctance of a material. R where RADIO Permeability Figure 20 TYPICAL HYSTERESIS LOOP (B -H CURVE = A -B) Showing relationship between the current in the winding of on iron core inductor and the core Inducflux. A direct current flowing through tance brings the magnetic state of the core to some point on the hysteresis loop, such as C. www.americanradiohistory.com th Inductance HANDBOOK original direction, the flux passes through a typical hysteresis loop as shown. The magnetism remaining in a material after the magnetizing force is removed is called residual magnetism. Retentivity is the property which causes a magnetic material to have residual magnetism after having been magnetized. Residual Magnetism; Retentivity Hysteresis; Coercive Force Hysteresis is the character - istic of a magnetic system which causes a loss of power due to the fact that a negative magnetizing force must be applied to reduce the residual magnetism to zero. This negative force is termed coercive /orce. By "negative" magnetizing force is meant one which is of the opposite polarity with respect to the original magnetizing force. Hysteresis loss is apparent in transformers and chokes by the heating of the core. the current. Thus, it can be seen that selfinduction tends to prevent any change in the current in the circuit. The storage of energy in a magnetic field is expressed in joules and is equal to (LI3) /2. (A joule is equal to 1 watt- second. L is defined immediately following.) Inductance is usually denoted by the letter L, and is expressed in henrys. A coil has an inductance of 1 henry when a voltage of 1 volt is induced by a current change of 1 ampere per second. The henry, while commonly used in audio frequency circuits, is too large for reference to inductance coils, such as those used in radio frequency circuits; millihenry or microhenry is more commonly used, in the following manner: The Unit of Inductance; The Henry 1 1 If the switch shown in figure 19 is opened and closed, a pulsating direct current will be produced. When it is first closed, the current does not instantaneously rise to its maximum value, but builds up to it. While it is building up, the magnetic field is expanding around the conductor. Of course, this happens in a small fraction of a second. If the switch is then opened. the current stops and the magnetic field contracts quickly. This expanding and contracting field will induce a current in any other conductor that is part of a continuous circuit which it cuts. Such a field can be obtained in the way just mentioned by means of a vibrator inter ruptor, or by applying a.c. to the circuit in place of the battery. Varying the resistance of the circuit will also produce the same effect. This inducing of a current in a conductor due to a varying current in another conductor not in acutal contact is called electromagnetic induction. Inductance Self -inductance If an alternating current flows through a coil the varying magnetic field around each turn cuts itself and the adjacent turn and induces a voltage in the coil of opposite polarity to the applied e.m.f. The amount of induced voltage depends upon the number of turns in the coil, the current flowing in the coil, and the number of lines of force threading the coil. The voltage so induced is known as a counter-e.m. f. or back e.m.f., and the effect is termed self -induction. When the applied voltage is building up, the counter- e.m.f. opposes the rise; when the applied voltage is decreasing, the counter- e.m.f. is of the same polarity and tends to maintain 37 henry = 1,000 henrys. 1 or 10' milli - millihenry = 1 /1,000 of a henry, .001 henry, or 1 millihenrys, 10' henry. microhenry = 1 /1,000,000 of a henry, .000001 henry, or 10-e henry. or microhenry =1/1,000 of a millihenry, .001 or 10-' millibenrys. 1,000 microbenrys = millihenry. 1 coil is near another, a varying current in one will produce a varying magnetic field which cuts the turns of the other coil, inducing a current in it. This induced current is also varying, and will therefore induce another current in the first coil. This reaction between two coupled circuits is called mutual induction, and can be calculated and expressed in henrys. The symbol for mutual inductance is M. Two circuits thus joined are said to be inductively coupled. The magnitude of the mutual inductance depends upon the shape and size of the two circuits, their positions and distances apart, and the premeability of the medium. The extent to When one Mutual Inductance i I i., u I I 2 I Figure 21 MUTUAL INDUCTANCE The quantity M represents the mutual inductance between the two coils L1 and L,. www.americanradiohistory.com Direct Current Circuits 38 i-- L ---¡ INDUCTANCE OF SINGLE- LAYER SOLENOID COILS R2 N2 9R +10 L L WHERE R = L = RADIUS OF COIL LENGTH OF COIL N = NUMBER OF TURNS MICRONENRIES TO CENTER OF WIRE Figure 22 FORMULA FOR CALCULATING INDUCTANCE Through the usa of the equation and the sketch shown above inductance of single -layer solenoid coils can be calculated with an accuracy of about on. per cent for tho types of coils normally used in the h -f and v -h -f range. th which two inductors are coupled is expressed by a relation known as coefficient of coupling. This is the ratio of the mutual inductance actually present to the maximum possible value. The formula for mutual inductance is L L, + L, + 2M when the coils are poled so that their fields add. When they are poled so that their fields buck, then L = L, + L, - 2M (figure 21). Inductors in parallel are corn bined exactly as are resistors in parallel, provided that they are far enough apart so that the mutual inductance is entirely negligible. Inductors in Parallel Inductors in series are additive, just as are resistors in series, again provided that no mutual inductance exists. In this case, the total inductance L is: Inductors in Series L = L, etc. + L2 + Where mutual inductance does exist: L =L, +L,+ M is the mutual inductance. This latter expression assumes that the coils are connected in such a way that all flux linkages are in the same direction, i.e., additive. If this is not the case and the mutual linkages subtract from the self -linkages, the following formula holds: L M =L, +L,- 2M, as the frequency is increased. The principal use for conventional magnetic cores is in the audio -frequency range below approximately 15,000 cycles, whereas at very low frequencies (50 to 60 cycles) their use is mandatory if an appreciable value of inductance is desired. An air core inductor of only 1 henry inductance would be quite large in size, yet values as high as 500 henrys are commonly available in small iron core chokes. The inductance of a coil with a magnetic core will vary with the amount of current (both a-c and d-c) which passes through the coil. For this reason, iron core chokes that are used in power supplies have a certain inductance rating at a predetermined value of d-c. The premeability of air does not change with flux density; so the inductance of iron core coils often is made less dependent upon flux density by making part of the magnetic path air, instead of utilizing a closed loop of iron. This incorporation of an air gap is necessary in many applications of iron core coils, particularly where the coil carries a considerable d -c component. Because the permeability of air is so much lower than that of iron, the air gap need comprise only a small fraction of the magnetic circuit in order to provide a substantial proportion of the total reluctance. Iron Cored Inductors at Radio Frequencies Iron -core inductors may be used at radio frequencies if the iron is in a very finely divided form, as in the case of the powdered iron cores used in some types of r -f coils and i -f transformers. These cores are made of extremely small particles of iron. The particles are treated with an insulating material so that each particle will be insulated from the others, and the treated powder is molded with a binder into cores. Eddy current losses are greatly reduced, with the result that these special iron cores are entirely practical in circuits which operate up to 100 Mc. in frequency. 2 -5 and R L R C voltage divider may be constructed as figure 23. Kirchhoff's and Ohm's Laws hold for such a divider. This circuit is known as an RC circuit. A - Circuits Ordinary magnetic cores cannot be used for radio frequencies because the eddy current and hysteresis losses in the core material becomes enormous Transients shown in Time Constant RC and RL is the mutual inductance. Core Material RADIO 2M, where where THE When switch S in figure 23 is placed in position 1, a volt meter across capacitor C will indicate the manner in which the capacitor will become charged through the resistor R from battery B. If relatively large values are used for R and C, and if a v -t voltmeter which draws negligible current is used www.americanradiohistory.com HANDBOOK Time Constant 39 to measure the voltage e, the rate of charge of the capacitor may actually be plotted with the aid of a stop watch. It will be found that the voltage e will begin to rise rapidly from zero the instant the switch is closed. Then, as the capacitor begins to charge, the rate of change of voltage across the capacitor will be found to decrease, the charging taking place more and more slowly as the capacitor voltage e approaches the battery voltage E. Actually, it will be found that in any given interval a constant percentage of the remaining difference between e and E will be delivered to the capacitor as an increase in voltage. A voltage which changes in this manner is said to increase logarithmically, or is said to follow an exponential curve. Voltage Gradient t001 r : 60 L< 60 W V40 < ti Wo rt 20 OTIME 44 100 <` < FaA 60 óóZ t. IN TERMS OF TIME CONSTANT PC' 030 60 Hi Iug 4.9 40. ózó 0 óZW <7 á 7ló1 ñ51 20¡C - -- 0 TIME Time Constant - t, IN TERMS OF TIME Figure 2 A mathematical analysis of the charging of a capacitor in this manner would show that the relationship between the battery voltage E and the voltage across the capacitor e could be expressed in the following manner: 13 CONSTANT RC 23 TIME CONSTANT OF AN R -C CIRCUIT Shown at (A) is the circuit upon which is based the curves of (B) and (C). (8) shows the rate at which capacitor C will charge from the instant at which switch S is placed in position 1. (C) shows the discharge curve of capacitor C from the instant at which switch S is placed in position 3. e = E (1 _ f -t /Rc) where e,E,R, and C have the values discussed above. f = 2.716 (the base of Naperian or natural logarithms), and t represents the time which has elapsed since the closing of the switch. With t expressed in seconds, R and C Figure 24 TYPICAL INDUCTANCES The large inductance is a 1000 -watt transmitting coil. To the right and left of this coil are small r -f chokes. S I varieties of low power capability coils are shown below, along with various types of r -f chokes intended for high- frequency operation. www.americanradiohistory.com Direct Current Circuits 40 R (INCLUDING D.C. RESISTANCE OF INDUCTOR L) i4cc means that the voltage across the capacitor will have increased to 63.2 per cent of the battery voltage in an interval equal to the time constant or RC product of the circuit. Then, during the next period equal to the time constant of the RC combination, the voltage across the capacitor will have risen to 63.2 per cent of the remaining difference in voltage, or 86.5 per cent of the applied voltage E. RL Circuit TIME t, IN TERMS OF TIME CONSTANT } Figure 25 In the case of a series combination of a resistor and an inductor, as shown in figure 25, the current through the combination follows a very similar law to that given above for the voltage appearing across the capacitor in an RC series circuit. The equation for the current through the combination is: TIME CONSTANT OF AN R -L CIRCUIT Nota that the time constant for the Increase In current through an R-L. circuit Is identical to the rate of Increase in voltage across the capacitor In on R -C circuit. may be expressed in farads and ohms, or R and C may be expressed in microfarads and megohms. The product RC is called the time constant of the circuit, and is expressed in seconds. As an example, if R is one megohm and C is one microfarad, the time constant RC will be equal to the product of the two, or one second. When the elapsed time t is equal to the time constant of the RC network under consideration, the exponent of E becomes -1. Now is equal to 1 /e, or 1/2.716, which is 0.368. The quantity (1 - 0.368) then is equal to 0.632. Expressed as percentage, the above e' i=-E (1-E-tR/L) where i represents the current at any instant through the series circuit, E represents the applied voltage, and R represents the total resistance of the resistor and the d-c resistance of the inductor in series. Thus the time constant of the RL circuit is L /R, with R expressed in ohms and L expressed in henrys. Voltage Decoy When the switch in figure 23 is moved to position 3 after the capacitor has been charged, the capacitor voltage will drop in the manner shown in figure 23 -C. In this case the voltage across the capacitor will decrease to 36.8 per cent of the initial voltage (will make 63.2 per cent of the total drop) in a period of time equal to the time constant of the RC circuit. TYPICAL IRON -CORE INDUCTANCES At the right is an upright mounting filter choke intended for use in low powered transmitters and audio equipment. At the center is o hermetically sealed inductance for use under poor environmental conditions. To the left is an inexpensive receiving -type choke, with a small iron -core r -f choke directly in front of it. www.americanradiohistory.com CHAPTER THREE Alternating Current Circuits present the usable frequency range for alternating electrical currents extends over the enormous frequency range from about 15 cycles per second to perhaps 30,000,000,000 cycles per second. It is obviously cumbersome to use a frequency designation in c.p.s. for enormously high frequencies, so three common units which are multiples of one cycle per second have been established. The previous chapter has been devoted to discussion of circuits and circuit elements upon which is impressed a current consisting of a flow of electrons in one direction. This type of unidirectional current flow is called direct current, abbreviated d. c. Equally as important in radio and communications work, and power practice, is a type of current flow whose direction of electron flow reverses periodically. The reversal of flow may take place at a low rate, in the case of power systems, or it may take place millions of times per second in the case of communications frequencies. This type of current flow is called alternating current, abbreviated a. c. At Frequency Spectrum a z 4- Y.1 ¢ K U 3 -1 TIME-41. a DIRECT CURRENT Alternating Current t CYCLE Frequency of on Alternating Current An -i alternating current is one whose amplitude of current flow periodically rises from zero to a maximum in one direction, decreases to zero, changes its direction, rises to maximum in the opposite direction, and decreases to zero again. This complete process, starting from zero, passing through two maximums in opposite directions, and returning to zero again, is called a cycle. The number of times per second that a current passes through the complete cycle is called the frequency of the current. One and one quarter cycles of an alternating current wave are illustrated diagrammatically in figure 1. Iz w CYCLE -01 TIME a CC J U ALTERNATING CURRENT Figure 1 ALTERNATING CURRENT AND DIRECT CURRENT Graphical comparison between unidrectionai (direct) current and alternating current as plotted against time. 41 www.americanradiohistory.com - 42 Alternating Current Circuits THE RADIO These units are: (1) the kilocycle (abbr., kc.), 1000 c.p.s. (2) the Megacycle (abbr., Mc.), 1,000,000 c.p.s. or 1000 kc. (3) the kilo -Megacycle (abbr., kN1c.), 1,000,000,000 c.p.s. or 1000 Mc. easily handled units such as these we can classify the entire usable frequency range into frequency bands. The frequencies falling between about 15 and 20,000 c.p.s. are called audio frequencies, abbreviated a.f., since these frequencies are audible to the human ear when converted from electrical to acoustical signals by a loudspeaker or headphone. Frequencies in the vicinity of 60 c.p.s. also are called power frequencies, since they are commonly used to distribute electrical power to the consumer. The frequencies falling between 10,000 c.p.s. (10 kc.) and 30,000,000.000 c.p.s. (30 kMc.) are commonly called radio frequencies, abbreviated r. J., since they are commonly used in radio communication and allied arts. The radio- frequency spectrum is often arbitrarily classified into seven frequency bands, each one of which is ten times as high in frequency as the one just below it in the spectrum (except for the v -1 -f band at the bottom end of the spectrum). The present spectrum, with classifications, is given below. With Frequency kc. 30 to 300 kc. 300 to 3000 kc. 3 to 30 Mc. 30 to 300 Mc. 300 to 3000 Mc. 3 to 30 kMc. 30 to 300 kMc. 10 to 30 Generation of Alternating Current Classification Very -low frequencies Low frequencies Medium frequencies High frequencies Very -high frequencies Ultra -high frequencies Abbrev. v.l.f l.f. m.f. h. f. v.h.f. u.h.f. Super-high frequencies s.h.f. Extremely -high frequencies e.h.f. Faraday discovered that if a conductor which forms part of a closed circuit is moved through a magnetic field so as to cut across the lines of force, a current will flow in the conductor. He also discovered that, if a conductor in a second closed circuit is brought near the first conductor and the current in the first one is varied, a current will flow in the second conductor. This effect is known as induction, and the currents so generated are induced currents. In the latter case it is the lines of force which are moving and cutting the second conductor, due to the varying current strength in the first conductor. A current is induced in a conductor if there is a relative motion between the conductor and a magnetic field, its direction of flow depending upon the direction of the relative Figure 2 THE ALTERNATOR Semi -schematic representation of the simplest form of the alternator. motion between the conductor and the field, and its strength depends upon the intensity of the field, the rate of cutting lines of force, and the number of turns in the conductor. machine that generates an alternating current is called an alternator or a -c generator. Such a machine in its basic form is shown in figure 2. It consists of two permanent magnets, M. the opposite poles of which face each other and are machined so that they have a common radius. Between these two poles, north (N) and south (S), a substantially constant magnetic field exists. If a conductor in the form of C is suspended so that it can be freely rotated between the two poles, and if the opposite ends of conductor C are brought to collector rings, there will be a flow of alternating current when conductor C is rotated. This current will flow out through the collector rings R and brushes B to the external circuit, X -Y. The field intensity between the two pole pieces is substantially constant over the entire area of the pole face However, when the conductor is moving parallel to the lines of force at the top or bottom of the pole faces, no lines are being cut. As the conductor moves on across the pole face it cuts more and more lines of force for each unit distance of travel, until it is cutting the maximum number of lines when opposite the center of the pole. Therefore, zero current is induced in the conductor at the instant it is midway between the two poles, and maximum current is induced when it is opposite the center of the pole face. After the conductor has rotated through 180° it can be seen that its position with respect to the pole pieces will be exactly opposite to that when it started. Hence, the second 180° of rotation will produce an alternation of current in the opposite direction to that of the first alternation. The current does not increase directly as the angle of rotation, but rather as the sine of the angle; hence, such a current has the mathematical form of a sine wave. Although Alternators A www.americanradiohistory.com HANDBOOK LINES Sine /lave The 43 OF FORCE t CYCLE E le2 90 60 A B C O E CYCLE 30 HM- --+ CYCLE- w t20 ISO te0 3 a 2 LINES OF FORCE (UNIFORM DENSITY 240 2t0 3 CYCLE' Graph showing sine -wave output current of the alternator of figure CYCLE' +- WHERE F = TIMES 330 2143 t Figure 3 OUTPUT OF THE ALTERNATOR -- 300 FREQUENCY IN CYCLES 2. Figure most electrical machinery does not produce a strictly pure sine curve, the departures are usually so slight that the assumption can be regarded as fact for most practical purposes. All that has been said in the foregoing paragraphs concerning alternating current also is applicable to alternating voltage. The rotating arrow to the left in figure 3 represents a conductor rotating in a constant magnetic field of uniform density. The arrow also can be taken as a vector representing the strength of the magnetic field. This means that the length of the arrow is determined by the strength of the field (number of lines of force), which is constant. Now if the arrow is rotating at a constant rate (that is, with constant angular velocity), then the voltage developed across the conductor will be proportional to the rate at which it is cutting lines of force, which rate is proportional to the vertical distance between the tip of the arrow and the horizontal base line. If EO is taken as unity or a voltage of 1, then the voltage (vertical distance from tip of arrow to the horizontal base line) at point C for instance may be determined simply by referring to a table of sines and looking up the sine of the angle which the arrow makes with the horizontal. When the arrow has traveled from A to point E, it has traveled 90 degrees or one quarter cycle. The other three quadrants are not shown because their complementary or mirror relationship to the first quadrant is obvious. It is important to note that time units are represented by degrees or quadrants. The fact that AB, BC, CD, and DE are equal chords (forming equal quadrants) simply means that the arrow (conductor or vector) is traveling at a constant speed, because these points on the radius represent the passage of equal units of time. The whole picture can be represented in another way, and its derivation from the foregoing is shown in figure 3. The time base is represented by a straight line rather than by 4 THE SINE WAVE illustrating one cycle of o sine wave. One complete cycle of alternation is broken up into 360 degrees. Then one -half cycle is 180 degrees, one -quarter cycle is 90 degrees, and so on down to the smallest division of the wave. A cosine wave has a shape identical to a sine wave but is shifted 90 degrees in phase In other words the wove begfna at full am plilude, the 90- degree point comes at zero amplitude, the 180 -degree point comes at full amplitude in the opposite direction of current How, etc. - angular rotation. Points A, B, C, etc., represent the same units of time as before. When the voltage corresponding to each point is projected to the corresponding time unit, the familiar sine curve is the result. The frequency of the generated voltage is proportional to the speed of rotation of the alternator, and to the number of magnetic poles in the field. Alternators may be built to produce radio frequencies up to 30 kilocycles, and some such machines are still used for low frequency communication purposes. By means of multiple windings, three -phase output may be obtained from large industrial alternators. From figure 1 we see that the value of an a -c wave varies continuously. It is often of importance to know the amplitude of the wave in terms of the total amplitude at any instant or at any time within the cycle. To be able to establish the instant in question we must be able to divide Radian Notation the cycle into parts. We could divide the cycle into eighths, hundredths, or any other ratio that suited our fancy. However, it is much more convenient mathematically to divide the cycle either into electrical degrees (360° represent one cycle) or into radians. A radian is an arc of a circle equal to the radius of the circle; hence there are 2n radians per cycle-or per circle (since there are n diameters per circumference, there are 2rr radii). Both radian notation and electrical degree www.americanradiohistory.com notation are used in discussions of alternating current circuits. However, trigonometric tables are much more readily available in terms of degrees than radians, so the following simple conversions are useful. 2n radians = 1 cycle = 3600 n radians ='/2 cycle = 180° - radians ='4 cycle RADIO THE Alternating Current Circuits 44 WHERE o e (THETA). PHASE B RADIANS = A A B'/r D. t ANGLE OR RADIANS OR T .277F T 90' IRO RADIANS OR 2 A RADIANS OR 270 350 RADIAN a 57.324 DEGREES n = 2 - radians ='4 cycle 90° Figure n = 60° = 45° 3 -4 radians =' n /R 1 radian cycle Current where e The instantaneous volt age or current is proportional to the sine of the angle through which the rotating vector has travelled since reference time t = 0. Hence, when the peak value of the a -c wave amplitude (either voltage or current amplitude) is known, and the angle through which the rotating vector has travelled is established, the amplitude of the wave at this instant can be determined through use of the following expression: e = Erna: sin matical relationships involving phase angles since such relationships are simplified when radian notation is used -cycle = 57.3° When the conductor in the simple alternator of figure 2 has made one complete revolution it has generated one cycle and has rotated through 2n radians. The expression 2nf then represents the number of radians in one cycle multiplied by the number of cycles per second (the frequency) of the alternating voltage or current. The expression then represents the number of radians per second through which the conductor has rotated. Hence 27rf represents the angular velocity of the rotating conductor, or of the rotating vector which represents any alternating current or voltage, expressed in radians per second. In technical literature the expression 2nf is often replaced by al, the lower -case Greek letter omega. Velocity multiplied by time gives the distance travelled. so 2nft (or an) represents the angular distance through which the rotating conductor or the rotating vector has travelled since the reference time t = 0. In the case of a sine wave the reference time t = 0 represents that instant when the voltage or the current, whichever is under discussion, also is equal to zero. Instantaneous Value The radian is a unit of phase angle, equal to 57.324 degrees. It is commonly used in mathe- 1 = 2n of Voltage or 5 ILLUSTRATING RADIAN NOTATION left, = the instantaneous voltage crest value of voltage, E = maximum f = frequency in cycles per second, and has elapsed expressed as a fraction of one second. t = period of time which since t = 0 The instantaneous current can be found from the same expression by substituting i for e and Imax for Emaz. It is often easier to visualize the process of determining the instantaneous amplitude by ignoring the frequency and considering only one cycle of the a -c wave. In this case, for a sine wave, the expression becomes: e=Ema: sin 9 where O represents the angle through which the vector has rotated since time (and amplitude) were zero. As examples: when 0 = 30° sin O = 0.5 so e = 0.5 Emu when O = sin O so e = = 60° 0.866 0.866 Erna: when O = 90° sin 0 = 1.0 so e = Emax when sin so www.americanradiohistory.com O = = e = O 1 radian 0.8415 0.8415 Etna: H A-C A N D B O O K Effective Value The instantaneous of an of an alternating current or voltage varies continuously throughout the cycle. Alternating Current Relationships 45 value value of an a -c wave must be chosen to establish a relationship between the effectiveness of an a -c and a d -c voltage or cur rent/ The heating value of an alternating current has been chosen to establish the reference between the effective values of a.c. and d.c. Thus an alternating current will have an effective value of 1 ampere when it produces the same heat in a resistor as does 1 ampere of direct current. The effective value is derived by taking the instantaneous values of current over a cycle of alternating current, squaring these values. taking an average of the squares, and then taking the square root of the average. By this procedure, the effective value becomes known as the root mean square or r.m.s. value. This is the value that is read on a -c voltmeters and a -c ammeters. The r.m.s. value is 70.7 (for sine waves only) per cent of the peak or maximum instantaneous value and is expressed as So some follows: Eetf. or Er.m.s. = 0.707 x left. or Ir.m.s. = 0.707 x !ma:. Erna: or The following relations are extremely useful in radio and power work: Er m. s. = Ems 0.707 x = 1.414 x &max, and Er.m.s. If an alternating current is passed through a rectifier, it emerges in the form of a current of varying amplitude which flows in one direction only. Such a current is known as rectified a. c. or pulsating d. c. A typical wave form of a pulsating direct current as would be obtained from the output of a full -wave rectifier is shown in figure 6. Measuring instruments designed for d -c operation will not read the peak for instantaneous maximum value of the pulsating d -c output from the rectifier; they will read only the average value. This can be explained by assuming that it could be possible to cut off some of the peaks of the waves, using the cutoff portions to fill in the spaces that are open, thereby obtaining an average d -c value. A milliammeter and voltmeter connected to the adjoining circuit, or across the output of the rectifier, will read this average value. It is related to peak value by the following expres- Rectified Alternating Current or Pulsating Direct Current sion: Eavg = 0.636 x Fina: Figure 6 FULL -WAVE RECTIFIED SINE WAVE Waveform obtained at the output of a fullwave rectifier being fed with a sine wave and having 100 per cent rectification efficiency. Each pulse has the same shape os one -half cycle of a sine wave. This type of current is known as pulsating direct current. It is thus seen that the average value is 63.6 per cent of the peak value. To summarize the three most significant values of an a-c sine wave: the Effective, and Average Values peak value is equal to 1.41 times the r.m.s. or effective, and the r.m.s. value is equal to 0.707 times the peak value; the average value of a full -wave rectified a-c wave is 0.636 times the peak value, and the average value of a rectified wave is equal to 0.9 times the r.m.s. value. Relationship Between Peak, R.M.S. or = 0.707 x Peak Average = 0.636 x Peak R.M.S. x R.M.S. Average = 0.9 = 1.11 x Average R.M.S. Peak Peak = 1.414 x R.M.S. = 1.57 x Average law applies equally to direct or alternating current, provided the circuits under consideration are purely resistive, that is, circuits which have neither inductance (coils) nor capacitance (capacitors). Problems which involve tube filaments, drop resistors, electric lamps, heaters or similar resistive devices can be solved from Ohm's law, regardless of whether the current is direct or alternating. When a capacitor or coil is made a part of the circuit, a property common to either, called reactance, must be taken into consideration. Ohm's law still applies to a -c circuits containing reactance, but additional considerations are involved; these will be discussed in a later Applying Ohm's Law to Alternating Current paragraph. www.americanradiohistory.com Ohm's Alternating Current Circuits 46 THE RADIO E TIME TIME CURRENT LAGGING VOLTAGE BY 90° CURRENT LEADING VOLTAGE BY 90° (CIRCUIT CONTAINING PURE INDUCTANCE ONLY) (CIRCUIT CONTAINING PURE CAPACITANCE ONLY) Figure 7 LAGGING PHASE ANGLE Figure 8 LEADING PHASE ANGLE Showing the manner in which the current lags the voltage in an a-c circuit containing pure inductance only. The lag is equal to one -quarter cycle or 90 degrees. Inductive As Reactance when was stated in Chapter Two, changing current flows through an inductor a back- or counter -electromotive force is developed, opposing any change in the initial current. This property of an inductor causes it to offer opposition or impedance to a change in current. The measure of impedance offered by an inductor to an alternating current of a given frequency is known as its inductive reactance. This is expressed as XL. a XL = 2rrf L, n= 3.1416 (2n= 6.283), f = frequency in cycles, L = inductance in henrys. It is very often neces- cary to compute inductive reactance at radio frequencies. The same formula may be used, but to make it less cumbersome the inductance is expressed in millihenrys and the frequency in kilocycles. For higher frequencies and smaller values of inductance, frequency is expressed in megacycles and inductance in microhenrys. The basic equation need not be changed, since the multiplying factors for inductance and frequency appear in numerator and denominator, and hence are cancelled out. However, it is not possible in the same equation to express L in millihenrys and f in cycles without conversion factors. Capacitive Reactance Capacitors have a similar property although in this case the opposition is to any change in the voltage across the capacitor. This property is called capacitive reactance and is expressed as follows: Xc It has been explained that inductive reactance is the measure of the ability of an inductor to offer impedance to the flow of an alternating current. 1 2nfC where Xc = capacitive reactance in ohms, n = 3.1416 f = frequency in cycles, C = where XL = inductive reactance expressed in ohms. Inductive Reactance at Rodio Frequencies Showing the manner in which the current leads the voltage in on o -c circuit containing pure capacitance only. The lead is equal to one quarter cycle or 90 degrees. capacitance in farads. Capacitive Reactance at Radio Frequencies Here again, as in the case of inductive reactance, the units of capacitance and frequency can be converted into smaller units for practical problems en- countered in radio work. The equation may be written: 1,000,000 Xc 2nfC where f = frequency in megacycles, C w capacitance in micro- microfarads. In the audio range it is often convenient to express frequency (f) in cycles and capacitance (C) in micro /arads, in which event the same formula applies. Phase When an alternating current flows through a purely resistive circuit, it will be found that the current will go through maximum and minimum in perfect step with the voltage. In this case the current is said to be in step or in phase with the voltage. For this reason, Ohm's law will apply equally well for a. c. or d. c. where pure resistances are concerned, provided that the same values of the www.americanradiohistory.com Reactance HANDBOOK Y-AniS wave (either peak or r.m.s.) for both voltage and current are used in the calculations. However, in calculations involving alternating currents the voltage and current are not necessarily in phase. The current through the circuit may lag behind the voltage, in which case the current is said to have lagging phase. Lagging phase is caused by inductive reactance. If the current reaches its maximum value ahead of the voltage (figure 8) the current is said to have a leading phase. A leading phase angle is caused by capacitive reactance. In an electrical circuit containing reactance only, the current will either lead or lag the voltage by 90 °. If the circuit contains inductive reactance only, the current will lag the voltage by 90 °. If only capacitive reactance is in the circuit, the current will lead the voltage by 90 °. Inductive and capacitive reactance have exactly opposite effects on the phase relation between current and voltage in a circuit. Hence when they are used in combination their effects tend to neutralize. The combined effect of a capacitive and an inductive reactance is often called the net reactance of a circuit. The net reactance (X) is found by subtracting the capacitive reactance from the inductive reactance, X = XL Xc. The result of such a combination of pure reactances may be either positive, in which case the positive reactance is greater so that the net reactance is inductive, or it may be negative in which case the capacitive reactance is greater so that the net reactance is capacitive. The net reactance may also be zero in which case the circuit is said to be resonant. The condition of resonance will be discussed in a later section. Note that inductive reactance is always taken as being positive while capacitive reactance is always taken as being negative. Reactances in Combination Pure reactances introduce a phase angle of 90° between voltage and and Resistance current; pure resistance introduces no phase shift between voltage and current. Hence we cannot add a reactance and a resistance directly. When a reactance and a resistance are used in combination the resulting phase angle of current flow with respect to the impressed voltage lies somewhere between plus or minus 90° and 0° depending upon the relative magnitudes of the reactance and the resistance. The term impedance is a general term which can be applied to any electrical entity which impedes the flow of current. Hence the term may be used to designate a resistance, a pure Impedance; Circuits Containing Reactance 47 Figure 9 Operation on the vector (+A) by the quantity ( -1) vector to rotate through 180 degrees. reactance, or a complex combination of both reactance and resistance. The designation for impedance is Z. An impedance must be defined in such a manner that both its magnitude and its phase angle are established. The designation may be accomplished in either of two ways-one of which is convertible into the other by simple mathematical operations. "J" The first method of designating an impedance is actually to specify both the resistive and the reactive component in the form R + jX. In this form R represents the resistive component in ohms and X represents the reactive component. The "j" merely means that the X component is reactive and thus cannot be added directly to the R component. Plus jX means that the reactance is positive or inductive, while if minus jX were given it would mean that the reactive component was negative or capacitive. In figure 9 we have a vector ( +A) lying along the positive X-axis of the usual X -Y coordinate system. If this vector is multiplied by the quantity ( -1), it becomes ( A) and its position now lies along the X -axis in the negative direction. The operator ( -1) has caused the vector to rotate through an angle of 180 deThe Operator grees. Since ( -1) is equal to (V-1 x V77-1), the same result may be obtained by operating on the vector with the operator (VIT x V-1). However if the vector is operated on but once by the operator (V-1), it is caused to rotate only 90 degrees (figure 10). Thus the operator ( V- 11)rotates a vector by 90 degrees. For convenience, this operator is called the j operator. rotates the In like fashion, the operator ( vector of figure 9 through an angle of 270 degrees, so that the resulting vector ( jA) falls on the ( Y) axis of the coordinate system. www.americanradiohistory.com j) 48 Alternating Current Circuits RADIO THE Y-AXIS (AI ( +A) X ) ROTATES VECTOR THROUGH 90 +jA k 4 b A X Z. 4+J3 AXIS X i Figure (j) Polar Notation The second method of representing an impedance is to specify its absolute magnitude and the phase angle of current with respect to voltage, in the form Z L O. Figure 11 shows graphically the relationship between the two common ways of representing an impedance. The construction of figure 11 is called an impedance diagram. Through the use of such a diagram we can add graphically a resistance and a reactance to obtain a value for the resulting impedance in the scalar form. With zero at the origin, resistances are plotted to the right, positive values of reactance (inductive) in the upward direction, and negative values of reactance (capacitive) in the downward direction. Note that the resistance and reactance are drawn as the two sides of a right triangle, with the hypotenuse representing the resulting impedance. Hence it is possible to determine mathematically the value of a resultant impedance through the familiar right -triangle relationship-the square of the hypotenuse is equal to the sum of the squares of the other two sides: =R' +X2 or IZI = ß/R2 + X Note also that the angle O included between R and Z can be determined from any of the following trigonometric relationships: X sin o cos tan Z R O = Z X O = - IZI' -R OHMS RESISTANCE R. 5 IZI= 5 t0/-' 0.73 ae.e5 10 Operation on the vector ( +A) by the quantity causes vector to rotate through 90 degrees. Z2 o L 3e.e5 R One common problem is that of determining the scalar magnitude of the impedance, IZI, Figure 11 THE IMPEDANCE TRIANGLE Showing the graphical construction of a triangle for obtaining the net (scalar) impedance resulting from the connection of o resistance and a reactance in series. Shown also alongside is the alternative mathematical procedure for obtaining the values associated with the triangle. and the phase angle 0, when resistance and reactance are known; hence, of converting from the Z = R + jX to the IZI LO form. In this case we use two of the expressions just given: IZI = V/R2+X2 tan -, (or X O = O = R The tan' X ) R inverse problem, that of converting jX form is done with the following relationships, both of which are obtainable by simple division from the trigonometric expressions just given for determining the angle 0: from the IZI LO to the R + R jX =IZI cos0 =IZIj sin O By simple addition these two expressions may be combined to give the relationship between the two most common methods of indicating an impedance: R + jX =IZI (cos B + j sin 0) In the case of impedance, resistance, or reactance, the unit of measurement is the ohm; hence, the ohm may be thought of as a unit of opposition to current flow, without reference to the relative phase angle between the applied voltage and the current which flows. Further, since both capacitive and inductive reactance are functions of frequency, impedance will vary with frequency. Figure 12 shows the manner in which IZI will vary with frequency in an RL series circuit and in an RC series circuit. Series RLC Circuits www.americanradiohistory.com In a series circuit containing R, L, and C, the im- HANDBOOK Impedance 49 pedance is determined as discussed before except that the reactive component in the expressions becomes: (The net reactance-the difference between XL and Xc.) Hence (XL Xc) may be substituted for X in the equations. Thus: IZI = VR' +(XL O = tan (XL '(XL Xc)' Xc ) R A series RLC circuit thus may present an impedance which is capacitively reactive if the net reactance is capacitive, inductively reactive if the net reactance is inductive, or resistive if the capacitive and inductive reactances are equal. Addition of Complex Quantities The addition of complex quantities (for example, impedances in series) is quite simple if the quantities are in the rectangular form. If they are in the polar form they only can be added graphically, unless they are converted to the rectangular form by the relationships previously given. As an example of the addition of complex quantities in the rectangular form, the equation for the addition of impedances is: ( R, +jX,) +(R, +jX,)= (R, + Rs) +j(X,+X,) For example if we wish to add the impedances (10 + j50) and (20 j30) we obtain: (10 + j50) + (20 j30) = (10 + 20) + j(50 + ( -30) = 30 + j(50 -30) It is often necessary in solving certain types of circuits to multiply or divide two complex quantities. It is a much simplier mathematical operation to multiply or divide complex quantities if they are expressed in the polar form. Hence if they are given in the rectangular form they should be converted to the polar form before multiplication or division is begun. Then the multiplication is accomplished by multiplying the IZ1 terms together and adding algebraically the L e terms, as: (IZ,1Le,)(I4,I Le,) =IZ,1 Iz,I L43°) (1321 L -23 °) approaches zero. Division is accomplished by dividing the denominator into the numerator, and subtracting the angle of the denominator from that of the numerator, as: IZ,I Le, IZ,IL0,- = 120.321 (L43° + L -23 °) = 640 L 20° IZ,I 1z21(Le, tee,) For example, suppose that an impedance of 1501 L 67° is to be divided by an impedance of 1101 L45°. Then: 1501 1101 L67° L45° 1501 = 151 1101 (L 22 °) Ohm's Law for Complex Quantities The simple form of Ohm's Law used for d -c circuits may be stated in a more general form for application to a -c circuits involving either complex quantities or simple resistive elements. The form is: E z (L0,+ L0,) For example, suppose that the two impedances 1201 L43 and 1321 L -23° are to be multiplied. Then: ( 1201 Figure 12 IMPEDANCE AGAINST FREQUENCY FOR R L AND R -C CIRCUITS The impedance of an R -C circuit approaches infinity os the frequency approaches zero (d.c.), while the impedance of o series R -L circuit approaches infinity as the frequency approaches infinity. The impedance of an R -C circuit approaches the impedance of the series resistor os the frequency approaches infinity, while the impedance of o series R -L circuit approaches the impedance of the resistor as the frequency ) = 30 + j20 Multiplication and Division of Complex Quantities o in which, in the general case, I, E, and Z are complex (vector) quantities. In the simple case where the impedance is a pure resistance with an a -c voltage applied, the equation simplifies to the familiar I = E /R. In any case the applied voltage may be expressed either as peak, r.m.s., or average; the resulting www.americanradiohistory.com Alternating Current Circuits 50 THE Since the applied voltage will be the reference for the currents and voltages within the circuit, we may define it as having a zero phase angle: E = 100 LO °. Then: 200 n. oo _ Figure 13 R -L -C CIRCUIT = SERIES current always will be in the same type of units as used to define the voltage. In the more general case vector algebra must be used to solve the equation. And, since either division or multiplication is involved, the complex quantities should be expressed in the polar form. As an example, take the case of the series circuit shown in figure 13 with 100 volts applied. The impedance of the series circuit can best be obtained first in the rectangular form, as: ;(l00-.300) = 200 200 + j(100 V200'+(-200)' = N/40,000 + = 100 LO ° 282 L -45° 40,000 This same current must flow through all three elements of the circuit, since they are in series and the current through one must already have passed through the other two. Hence the voltage drop across the resistor (whose phase angle of course is 0 °) is: E = The voltage drop across the inductive reactance is: E = I XL (0.354 L45 °) (100 L 90 °) = 35.4 L 135° volts Similarly, the voltage drop across the capacitive reactance is: E = - E = (0.354 1_45°) (300 /--90 °) = -200 X tan' -=tan' 200 R = = tan' -1 -45 °. = 282 L -45° Note that in a series circuit the resulting impedance takes the sign of the largest reactance in the series combination. Where a slide-rule is being used to make the computations, the impedance may be found without any addition or subtraction operations by finding the angle O first, and then using the trigonometric equation below for obtaining the impedance. Thus: O -tan' -1 =tan'-XR =tan' -200 200 = -45° R Then IZI = cos cos -45° O 200 IZI L0°) 70.8 L 45° volts E = I Xc 80,000 Therefore Z =1R E = (0.354 L 45 °) (200 = 282 f2 O= -0 .354 L0 °- ( -45 °) 0.354 L45° amperes. -j200 Now, to obtain the current we must convert this impedance to the polar form. IZI = RADIO 0.707 = 0.707 Note that the voltage drop across the capacitive reactance is greater than the supply voltage. This condition often occurs in a series RLC circuit, and is explained by the fact that the drop across the capacitive reactance is cancelled to a lesser or greater extent by the drop across the inductive reactance. It is often desirable in a problem such as the above to check the validity of the answer by adding vectorially the voltage drops across the components of the series circuit to make sure that they add up to the supply voltage or to use the terminology of Kirchhoff's Second Law, to make sure that the voltage drops across all elements of the circuit, including the source taken as negative, is equal to zero. In the general case of the addition of a number of voltage vectors in series it is best to resolve the voltages into their in -phase and out-of -phase components with respect to the supply voltage. Then these components may be added directly. Hence: - ER = = - 28212 106.2 L -45° = = 70.8L45° 70.8 ( cos 45° + j sin 45 °) 70.8 (0.707 + j0.707) 50 + j50 www.americanradiohistory.com HANDBOOK Vector Algebra 51 ao i DROP ACROSS RESISTOR ,o °'- ±leo 45 e LI NE VOLTAGE =100 PARALLEL CIRCUIT XL + XL .70.111/-45. f0 Figure = = = Ec= 13. 35.4L135° 35.4 ( cos 135° + j sin 135 °) 35.4 ( -0.707 + j0.707) -25 + j25 106.2L45° = 106.2 ( cos -45 °+ j sin -45 °) = 106.2 (6.707 -j0.707) = 75 -j75 =(50+ j50) +(75 -j75) ER + EL +EC Figure + (-25 ments which go to make up the series circuit is the same. But the voltage drops across each of the components are, in general, different from one another. Conversely, in a parallel RLC or RX circuit the voltage is, obviously, the same across each of the elements. But the currents through each of the elements are usually different. There are many ways of solving a problem involving paralleled resistance and reactance; several of these ways will be described. In general, it may be said that the impedance of a number of elements in parallel is solved using the same relations as are used for solving resistors in parallel, except that complex quantities are employed. The basic re- lation is: + j25) 1 Zrot the supply voltage. It is frequently desirable to check computations in- volving complex quantities by constructing vectors representing the quantities on the complex plane. Figure 14 shows such a construction for the quantities of the problem just completed. Note that the answer to the problem may be checked by constructing a parallelogram with the voltage drop across the resistor as one side and the net voltage drop across the capacitor plus the inductor (these may be added algrebraically as they are 180° out of phase) as the adjacent side. The vector sum of these two voltages, which is represented by the diagonal of the parallelogram, is equal to the supply voltage of 100 volts at zero phase angle. Resistance and Reactonce in Parallel -+ -+ -+ Z, 1 -25+ = (50 75) + j(50 + 25-75) = 100 +j0 = 100 LO °, which is equal to Checking by Construction on the Complex Plane 15 THE EQUIVALENT SERIES CIRCUIT Showing a parallel R -C circuit and the equivalent series R -C circuit which represents the same net impedance os the parallel circuit. 14 Graphical construction of the voltage drops associated with the serles R -L -C circuit of figure EQUIVALENT SERIES CIRCUIT -43 1 NET DROP ACROSS ,.n - . DROP ACROSS XC =1Oe.2 EL = T :_e.i VOLTAGE DROP ACROSS X1.= 35.4 laa series circuit, such as just discussed, the current through all the ele- 1 1 Z2 Z, or when only two impedances are involved: Z, Z2 Z`o` Z t + Z : As an example, using the two- impedance relation, take the simple case, illustrated in figure 15, of a resistance of 6 ohms in parallel with a capacitive reactance of 4 ohms. To simplify the first step in the computation it is best to put the impedances in the polar form for the numerator, since multiplication is involved, and in the rectangular form for the addition in the denominator. Zrot - (6 L0°) (4 -90°) 6-j4 24 6 L-90° -j4 Then the denominator is changed to the polar form for the division operation: In a O = tañ' -4 www.americanradiohistory.com 6 = tan-' - 0.667 = - 33.7° THE Alternating Current Circuits 52 6 IZI = cos 7.21 ohms 0.832 33.7° E. j4 = 7.21 L-33.7° 6 ° Ztat = 7.21 = 3.33 L ( Rz EzEi R1+Rz L -90° 33.7 °= cos = 3.33 [ 0.5548 + j = 1.85 j 3.33 L -56.3° 56.3° + j sin (- 0.832)1 Through the series of operations in the previous paragraph we have converted a circuit composed of two impedances in parallel into an equivalent series circuit composed of impedances in series. An equivalent series circuit is one which, as far as the terminals are concerned, acts identically to the original parallel circuit; the current through the circuit and the power dissipation of the resistive elements are the same for a given voltage at the specified frequency. We can check the equivalent series circuit of figure 15 with respect to the original circuit by assuming that one volt a.c. (at the frequency where the capacitive reactance in the parallel circuit is 4 ohms) is applied to the terminals of both. In the parallel circuit the current through the resistor will be 2/6 ampere (0.166a.) while the current through the capacitor will be j ampere (+ j 0.25 a.). The total current will be the sum of these two currents, or 0.166 + j 0.25 a. Adding these vectorially we obtain: W 1 =0.3a. will be: =I2R =0.32x1.85 = 0.9 x 1.85 = 0.166 watts that the equivalent series circuit checks exactly with the original parallel circuit. So we see Parallel RLC Circuits In solving a more complicated circuit Ez-E G+Cz E2-E1 Lz LI+Lz Cz O elect to use either of two methods of solution. These methods are called the admittance method and the assumed - voltage method. However, the two methods are equivalent since both use the sum-of-reciprocals equation: may 1 Ztot - +- +1 1 1 Z1 Z2 Zs In the admittance method we use the relation Y = 1 /Z, where Y = G + jB; Y is called the admittance, defined above, G is the conductance or R /Z' and B is the susceptance or X/Z2. Then Ytot = 1 /Ztot = Y1 + Y2 + Y, In the assumed- voltage method we multiply both sides of the equation above by E, the assumed voltage, and add the currents, as: E Ztot -+ -+- ... E E E Zt Z, Z3 = 1zt +Iz2 +1zt .. Then the impedance of the parallel combination may be determined from the relation: Ztot = Eí IZ tot Voltage dividers for use with alternating current are quite similar to d-c voltage dividers. However, since capacitors and inductors oppose the flow of a-c current as well as resistors, voltage dividers for alternating voltages may take any of the configurations shown in figDividers The dissipation in the resistor will be 12/6 = 0.166 watts. In the case of the equivalent series circuit the current will be: And the dissipation in the resistor xo +502 AC Voltage III = x/0.1662 + 0.252 = x/0.09 = 0.3 a. 3.33 xCz Ez=E1 Figure 16 SIMPLE A -C VOLTAGE DIVIDERS Equivalent Series Circuit E +o OA 56.3°) 2.77 Ill ,o Ez 1 Then: 24 I 6 RADIO made up of more than two impedances in parallel we ure 16. Since the impedances within each divider are of the same type, the output voltage is in phase with the input voltage. By using com- binations of different types of impedances, the phase angle of the output may be shifted in relation to the input phase angle at the same time the amplitude is reduced. Several dividers of this type are shown in figure 17. Note that the ratio of output voltage to input voltage is equal to the ratio of the output impedance to the total divider impedance. This relationship is true only if negligible current is drawn by a load on the output terminals. www.americanradiohistory.com HANDBOOK Xc E2Ei Circuits Resonant E2 R2+XC2 E 53 XL R2 +XL2 Figure © 18 SERIES RESONANT CIRCUIT If the values of inductance and capacitance both are fixed, there will be only one resonant E2E, XL Ez XL-Xc DO E, Ei Ea Es - Ei R2+2 Xc - X R2.2 XL-XCXC R2+I (L-XC12 COMPLEX 3 -2 Figure 17 VOLTAGE DIVIDERS A -C Resonant Circuits frequency. If both the inductance and capacitance are made variable, the circuit may then be changed or tuned, so that a number of combinations of inductance and capacitance can resonate at the same frequency. This can be more easily understood when one considers that inductive reactance and capacitive reactance travel in opposite directions as the frequency is changed. For example, if the frequency were to remain constant and the values of inductance and capacitance were then changed, the following combinations would have equal reactance: Frequency is constant at 60 cycles. L is expressed in henrys. series circuit such as shown in figure 18 is said to be in resonance when the applied frequency is such that the capacitive reactance is exactly balanced by the inductive reactance. At this frequency the two reactances will cancel in their effects, and the impedance of the circuit will be at a minimum so that maximum current will flow. In fact, as shown in figure 19 the net impedance of a series circuit at resonance is equal to the resistance which remains in the circuit after the reactances have been cancelled. A resistance is always present in a circuit because it is possessed in some degree by both the inductor and the capacitor. If the frequency of the alternator E is varied from nearly zero to some high frequency, there will be one particular frequency at which the inductive reactance and capacitive reactance will be equal. This is known as the resonant frequency, and in a series circuit it is the frequency at which the circuit current will be a maximum. Such series resonant circuits are chiefly used when it is desirable to allow a certain frequency to pass through the circuit (low impedance to this frequency), while at the same time the circuit is made to offer considerable opposition to currents of other frequencies. R C is expressed in microfarads (.000001 farad.) XL L .265 2.65 26.5 265.00 2,650.00 1,000 10,000 100,000 1,000,000 Frequency of Resonance 100 1,000 10.000 100,000 1,000,000 From the formula for resonance, 2rrfL = 1 /2nfC. the resonant frequency is determined: f= t Frequency Some Xc C 26.5 2.65 .265 .0265 .00265 100 1 2rr N/ LC where f = frequency in cycles, L = inductance in henrys, C = capacitance in farads. It is more convenient to express L and C in smaller units, especially in making radio frequency calculations; f can also be expressed in megacycles or kilocycles. A very useful group of such formulas is: f2= 25,330 LC orL= 25,330 f2C orC= 25,330 f2L where f = frequency in megacycles, L = inductance in microhenrys, C = capacitance in micromicrofarads. www.americanradiohistory.com 54 Alternating Current Circuits THE Figure 19 IMPEDANCE OF A SERIES -RESONANT CIRCUIT Showing the variation in reactance of the separate elements and In the net impedance of o series resonant circuit (such as figure 18) with changing frequency. The vertical line is drawn at the point of resonance (XL Xc = 0) in the - series circuit. Impedance of Series Resonant Circuits 18) is: The impedance across the terminals of a series resonant circuit (figure /r3 Z = Xc)2, + (XL where Z = impedance in ohms, r = resistance in ohms, Xc = capacitive reactance in ohms, XL = inductive reactance in ohms. From this equation, it can be seen that the impedance is equal to the vector sum of the circuit resistance and the difference between the two reactances. Since at the resonant frequency XL equals Xc. the difference between them (figure 19) is zero, so that at resonance the impedance is simply equal to the resistance of the circuit; therefore, because the resistance of most normal radio- frequency circuits is of a very low order, the impedance is also low. At frequencies higher and lower than the resonant frequency, the difference between the reactances will be a definite quantity and will add with the resistance to make the impedance higher and higher as the circuit is tuned off the resonant frequency. If Xc should be greater than XL, then the term (XL Xc) will give a negative number. However, when the difference is squared the product is always positive. This means that the smaller reactance is subtracted from the larger, regardless of whether it be capacitive or inductive, and the difference squared. RADIO FREQUENCY Figure 20 RESONANCE CURVE Showing the increase in impedance at resonance for o parallel- resonant circuit, and similarly, the increase in current at resonance for a series- rsonant circuit. The sharpness of resonance is determined by the Q of the circuit, as illustrated by a comparison between A, B, and C. Current and Voltage in Series Resonant Formulas for calculating currents and voltages in a series resonant circuit are similar to those of Circuits Ohm's law. =-Z E I E = IZ The complete equations: E I V' r= + E = 1 (XL + (XL Xc)2 Xc)' Inspection of the above formulas will show the following to apply to series resonant circuits: When the impedance is low, the current will be high; conversely, when the impedance is high, the current will be low. Since it is known that the impedance will be very low at the resonant frequency, it follows that the current will be a maximum at this point. If a graph is plotted of the current against the frequency either side of resonance, the resultant curve becomes what is known as a resonance curve. Such a curve is shown in figure 20, the frequency being plotted against current in the series resonant circuit. Several factors will have an effect on the shape of this resonance curve, of which re- www.americanradiohistory.com Circuit HANDBOOK sistance and L -to -C ratio are the important considerations. The curves B and C in figure 20 show the effect of adding increasing values of resistance to the circuit. It will be seen that the peaks become less and less prominent as the resistance is increased; thus, it can be said that the selectivity of the circuit is thereby decreased. Selectivity in this case can be defined as the ability of a circuit to discriminate against frequencies adjacent to the resonant frequency. Because the a.c. or r -f voltage across a coil and capacitor is proportional to the reactance (for a given current), the actual voltages across the coil and across the capacitor may be many times greater than the terminal voltage of the circuit. At resonance, the voltage across the coil (or the capacitor) is Q times the applied voltage. Since the Q (or merit factor) of a series circuit can be in the neighborhood of 100 or more, the voltage across the capacitor, for example, may be high enough to cause flashover, even though the applied voltage is of a value considerably below that at which the capacitor is rated. Voltage Across Coil and Capacitor in Series Circuit -Sharp- extremely important property of a capacitor or an inductor is its factor of- merit, more generally called its Q. It is this factor, Q, which primarily determines the sharpness of resonance of a tuned circuit. This factor can be expressed as the ratio of the reactance to the resistance, as follows: Circuit Q An ness of Resonance Q- R The actual resistance in a wire or an inductor can be far greater than the d-c value when the coil is used in a radio -frequency circuit; this is because the current does not travel through the entire cross -section of the conductor, but has a tendency to travel closer and closer to the surface of the wire as the frequency is increased. This is known as the skin effect. The actual current -carrying portion of the wire is decreased, as a result of the skin effect, so that the ratio of a -c to d -c resistance of the wire, called the resistance ratio, is increased. The resistance ratio of wires to be used at frequencies below about 500 kc. may be materially reduced through the use of Utz wire. Litz wire, of the type commonly used to wind the coils of 455 -kc. i -f transformers, may consist of 3 to 10 strands of insulated wire, about No. 40 in size, with the individual Skin Effect Examination of the equation for determining Q might give rise to the thought that even though the resistance of an inductor increases with frequency, the inductive reactance does likewise, so that the Q might be a constant. Actually, however, it works out in practice that the Q of an inductor will reach a relatively broad maximum at some particular frequency. Hence, coils normally are designed in such a manner that the peak in their curve of Q with frequency will occur at the normal operating frequency of the coil in the circuit for which it is designed. The Q of a capacitor ordinarily is much higher than that of the best coil. Therefore, it usually is the merit of the coil that limits the overall Q of the circuit. At audio frequencies the core losses in an iron -core inductor greatly reduce the Q from the value that would be obtained simply by dividing the reactance by the resistance. Obviously the core losses also represent circuit resistance, just as though the loss occurred in the wire itself. Variation of Q with Frequency circuits, parallel resonance (more correctly termed anti resonance) is more frequently encountered than series resonance; in fact, it is the basic foundation of receiver and transmitter circuit operation. A circuit is shown in figure 21. Parallel In radio Resonance "Tank" In this circuit, as contrasted with a circuit for series resonance, L (inductance) and C (capacitance) are connected in parallel, yet the combination can be considered to be in series with the remainder of the circuit. This combination of L and C, in conjunction with R, the resistance which is principally included in L, is sometimes called a tank circuit because it effectively functions as a storage tank when incorporated in vacuum tube circuits. Contrasted with series resonance, there are two kinds of current which must be considered in a parallel resonant circuit: (1) the line current, as read on the indicating meter M (2) the circulating current which flows within the parallel L -C -R portion of the circuit. See figure 21. At the resonant frequency, the line current (as read on the meter M,) will drop to a very low value although the circulating current in the L -C circuit may be quite large. It is interesting to note that the parallel resonant circuit acts in a distinctly opposite manner to that of a series resonant circuit, in which the Circuit where R = total resistance. 55 strands connected together only at the ends of the coils. The 2rrfL Q www.americanradiohistory.com 56 A!ternating Current Circuits THE RADIO plifier circuit, the impedance curve must have a sharp peak in order for the circuit to be selective. If the curve is broad- topped in shape, both the desired signal and the interfering signals at close proximity to resonance will give nearly equal voltages on the grid of the tube, and the circuit will then be nonselective; i.e., it will tune broadly. Figure 21 PARALLEL- RESONANT CIRCUIT The inductance L and capacitance C comprise the reactive elements of the parallel -resonant (anti -resonant) tank circuit, and the resistance R indicates the sum of the r -f resistance of the coil and capacitor, plus the resistance coupled into the circuit from the external load. In most coses the tuning capacitor has much lower r -f resistance than the coil and can therefore be ignored in comparison with the coil resistance and the coupled -in resistance. The instrument M1 indicates the "line current" which keeps the circuit in a state of oscillation current is the some as the fundamental component of the plate current of a Class C amplifier which might be feeding the tank circuit. The instrument M2 indicates the "tank current" which is equal to the line current multiplied by the operating Q of the tank circuit. -this current is at a maximum and the impedance is minimum at resonance. It is for this reason that in a parallel resonant circuit the principal consideration is one of impedance rather than current. It is also significant that the impedance curve for parallel circuits is very nearly identical to that of the current curve for series resonance. The impedance at resonance is expressed as: Z- (2trf L)2 R where Z = impedance in ohms, L = inductance in henrys, f = frequency in cycles, R = resistance in ohms. Or, impedance can be expressed as a function of Q as: Z= 2nfLQ, showing that the impedance of a circuit is directly proportional to its effective Q at resonance. The curves illustrated in figure 20 can be applied to parallel resonance. Reference to the curve will show that the effect of adding resistance to the circuit will result in both a broadening out and lowering of the peak of the curve. Since the voltage of the circuit is directly proportional to the impedance, and since it is this voltage that is applied to the grid of the vacuum tube in a detector or am- highest possible voltage can be developed across a parallel resonant circuit, the impedance of this circuit must be very high. The impedance will be greater with conventional coils of limited Q when the ratio of inductance-to-capacitance is great, that is, when L is large as compared with C. When the resistance of the circuit is In order that the Effect of L/C Ratio in Parallel Circuits very low. XL will equal XC at maximum impedance. There are innumerable ratios of L and C that will have equal reactance, at a given resonant frequency, exactly as in the case in a series resonant circuit. In practice, where a certain value of inductance is tuned by a variable capacitance over a fairly wide range in frequency, the L/C ratio will be small at the lowest frequency end and large at the high -frequency end. The circuit, therefore, will have unequal gain and selectivity at the two ends of the band of frequencies which is being tuned. Increasing the Q of the circuit (lowering the resistance) will obviously increase both the selectivity and gain. Circulating Tank The Q of a circuit has definite bearing on the circulating tank current at resonance. This tank current is very nearly the value of the line current multiplied by the effective circuit Q. For example: an r -f line current of 0.050 amperes, with a circuit Q of 100, will give a circulating tank current of approximately 5 amperes. From this it can be seen that both the inductor and the connecting wires in a circuit with a high Q must be of very low resistance, particularly in the case of high power transmitters, if heat losses are to be held to a minimum. Because the voltage across the tank at resonance is determined by the Q, it is possible to develop very high peak voltages across a high Q tank with but little line current. Current at Resonance Effect of Coupling on Impedance output circuit, the Q of the parallel coupling becomes (tighter) coupling www.americanradiohistory.com a If a parallel resonant circuit is coupled to another circuit, such as an antenna impedance and the effective circuit is decreased as the closer. The effect of closer is the same as though an Circuit Impedance HANDBOOK COUPLING LOP O vE N 4E011.14 COUPLING MEDI U4 0 LOOSE COUPLING HIGH 0 57 Z t Figure EFFECT OF COUPLING ON actual resistance were added in series with the parallel tank circuit. The resistance thus coupled into the tank circuit can be considered as being reflected from the output or load circuit to the driver circuit. The behavior of coupled circuits depends largely upon the amount of coupling, as shown in figure 22. The coupled current in the secondary circuit is small, varying with frequency, being maximum at the resonant frequency of the circuit. As the coupling is increased between the two circuits, the secondary resonance curve becomes broader and the resonant amplitude increases, until the reflected resistance is equal to the primary resistance. This point is called the critical coupling point. With greater coupling, the secondary resonance curve becomes broader and develops double resonance humps, which become more pronounced and farther apart in frequency as the coupling between the two circuits is increased. Tank Circuit Flywheel Effect When the plate circuit of a Class B or Class C operated tube is connected to a parallel resonant circuit tuned to the same frequency as the exciting voltage for the amplifier, the plate current serves to maintain this L/C circuit in a state of oscillation. The plate current is supplied in short pulses which do not begin to resemble a sine wave, even though the grid may be excited by a sine wave voltage. These spurts of plate current are converted into a sine wave in the plate tank circuit by virtue of the "Q" or "flywheel effect" of the tank. If a tank did not have some resistance losses, it would, when given a "kick" with a single pulse, continue to oscillate indefinitely. With a moderate amount of resistance or "fric- tion" in the circuit the tank will still have 22 CIRCUIT IMPEDANCE AND Q inertia, and continue to oscillate with decreasing amplitude for a time after being given a "kick." With such a circuit, almost pure sine -wave voltage will be developed across the tank circuit even though power is supplied to the tank in short pulses or spurts, so long as the spurts are evenly spaced with respect to time and have a frequency that is the same as the resonant frequency of the tank. Another way to visualize the action of the tank is to recall that a resonant tank with moderate Q will discriminate strongly against harmonics of the resonant frequency. The distorted plate current pulse in a Class C amplifier contains not only the fundamental frequency (that of the grid excitation voltage) but also higher harmonics. As the tank offers low impedance to the harmonics and high impedance to the fundamental (being resonant to a sinethe latter), only the fundamental appears across the tank circuit wave voltage in substantial magnitude. - - Confusion sometimes exists as to the relationship between the unloaded and the loaded Q of the tank circuit in the plate of an r -f power amplifier. In the normal case the loaded Q of the tank circuit is determined by such factors as the operating conditions of the amplifier, bandwidth of the signal to be emitted, permissible level of harmonic radiation, and such factors. The normal value of loaded Q for an r-f amplifier used for communications service is from perhaps 6 to 20. The unloaded Q of the tank circuit determines the efficiency of the output circuit and is determined by the losses in the tank coil, its leads and plugs and jacks if any, and by the losses in the tank capacitor which ordinarily are very low. The unloaded Q of a good quality large diameter tank coil in the high - frequency range may be as high as 500 Loaded and Unloaded 0 www.americanradiohistory.com THE Alternating Current Circuits 58 to 800, and values greater than 300 are quite -FUNDAMENTAL SINE WAVE(A - FUNDAMENTAL PLUS 3RD HARMONIC(C) common. -SQUARE WAVE Tank Circuit Since the unloaded Q of a tank circuit is determined by the minimum losses in the tank, while the loaded Q is determined by useful loading of the tank circuit from the external load in addition to the internal losses in the tank circuit, the relationship between the two Q values determines the operating efficiency of the tank circuit. Expressed in the form of an equation, the loaded efficiency of a tank Efficiency 3RD HARMONIC Figure circuit is: Tank efficiency = 1 _Qt x 100 3 -3 23 FUNDAMENTAL PLUS 3RD HARMONIC unloaded Q of the tank circuit Qi = loaded Q of the tank circuit As an example, if the unloaded Q of the tank circuit for a class C r -f power amplifier is 400, and the external load is coupled to the tank circuit by an amount such that the loaded Q is 20, the tank circuit efficiency will be: eff. = (1 - 20/400) x 100, or (1 - 0.05) x 100, or 95 per cent. Hence 5 per cent of the power output of the Class C amplifier will be lost as heat in the tank circuit and the remaining 95 per cent will be delivered to the load. Qu = FUNDAMENTAL PLUS 3RD AND 5TH HARMONICS(E) VI% HARMONIC (D) Figure 24 THIRD HARMONIC WAVE PLUS FIFTH HARMONIC FUNDAMENTAL PLUS 3RD. 5TH, AND 7TH HARMONICS FUNDAMENTAL PLUS 3RD AND STM HARMONICS SQUARE WAVE (G) Nonsinusoidal Waves and Transients 7TM HARMONIC Pure sine waves, basic wave shapes. and complex shape particularly square and peaked waves. (B) COMPOSITE WAVE-FUNDAMENTAL PLUS THIRD HARMONIC Qu where RADIO (F ) discussed previously, are Waves of many different are used in electronics, waves, saw -tooth waves, Any periodic wave (one that repeats itself in definite time intervals) is composed of sine waves of different frequencies and amplitudes, added together. The sine wave which has the same frequency as the complex, periodic wave is called the fundamental. The frequencies higher than the fundamental are called harmonics, and are always a whole number of times higher than the fundamental. For example, the frequency twice as high as the fundamental is called the second harmonic. Wave Composition The Square Wave Figure 23 compares a square wave with a sine wave (A) of the same frequency. If another sine wave (B) of smaller amplitude, but three times the frequency of (A), called the third harmonic, is added to (A), the resultant wave (C) more nearly approaches the desired square wave. Figure 25 RESULTANT WAVE, COMPOSED OF FUNDAMENTAL, THIRD, FIFTH, AND SEVENTH HARMONICS This resultant curve (figure 24) is added to fifth harmonic curve (D), and the sides of the resulting curve (E)are steeper than before. This new curve is shown in figure 25 after a 7th harmonic component has been added to it, making the sides of the composite wave even steeper. Addition of more higher odd harmonics will bring the resultant wave nearer and nearer a to the desired square wave shape. The square wave will be achieved if an infinite number of odd harmonics are added to the original sine wave. www.americanradiohistory.com Nonsinusoidal HANDBOOK FUND.PLUS 2ND ]RD, 4TH, NAR MENICS AND FUND PLUS 210 HARM. FUNDAMENTAL 2ND HARM. Waves 59 FUNDAMENTAL PLUS 3RD HARMONIC FUNDAMENTAL PLUS 2ND 3RD, iikkatIPUZTD"' TM NARMOrMCS ]TM HARMONIC 3RD HARMONIC -4111111 FVND. HARMONICS f YNO. PLU12Np3RD ATM\ STM AND STM NARMISN ICS FUND, PLUS AND AND PLUS END MARY. 3RD MARYON IC FUND. PLUS 2ND, 3RD ATM. \AND STM MARMONIOS T TH HARMONIC Z FUND. PLUS 2ND, 3RD, H XA RM ONI C4 S 2CS AN A M &IL" FUND PLUS 2040, 3R0,4T14, STM. TH, AND ?TM HARMS. DL S:w ÁNDU TMNthreCCaiTa CATM 7TH HARMONIC FUNDAMENTAL PLUS 3RD AND STD HARMONICS ,FUNDAMENTAL PLUS 3RD HARM. 5TH HARMONIC TOOTH WADE SA / FUND. PLUS 2ND, 3RD4 ATM, STM, ATM, AND 7TH HARMONICS Figure 26 COMPOSITION OF A SAWTOOTH WAVE FUNDAMENTAL PLUS 3RD, STN, AND 7TH HARMONICS FUNDAMENTAL PLUS 3RD ANO STH HARMONIC the same fashion, a sawtooth wave is made up of different sine waves (figure 26). The addition of all harmonics, odd and even, produces the sawtooth wave form. The Sawtooth Wave In 7TH HARMONIC / Figure 27 shows the composition of a peaked wave. Note how the addition of each sucessive harmonic makes the peak of the resultant higher and the sides steeper. The Pecked Wave The three preceeding examples show how a complex periodic wave is composed of a fundamental wave and different harmonics. The shape of the resultant wave depends upon the harmonics that are added, their relative amplitudes, and relative phase relationships. In general, the steeper the sides of the waveform, the more harmonics it contains. Figure 27 COMPOSITION OF A PEAKED WAVE Other Waveforms If an a -c voltage is substituted for the d-c input voltage in the RC Transient circuits discussed in Chapter 2, the same principles may be applied in the analysi* of the transient behavior. An RC coupling circuit is designed to have a long time constant with respect to the lowest frequency it must pass. Such a circuit is shown in figure 28. If a nonsinusoidal voltage is to be passed unchanged through the coupling circuit, the time constant AC Transient Circuits must be long with respect to the period of the lowest frequency contained in the voltage wave. An RC voltage divider that is designed to distort the input waveform is known as a differentiator or integrator, depending upon the locations of the output taps. The output from a differentiator is taken across the resistance, while the output from an integrator is taken across the capacitor. Such circuits will change the shape of any complex a-c waveform that is impressed upon them. This distortion is a function of the value of the time constant of the circuit as compared to the period of the waveform. Neither a differentiator nor an integrator can change the RC Differentiator and Integrator www.americanradiohistory.com 60 Alternating Current Circuits Cr 0.1 .Uf 100 v. 1000 C.P 5 R THE o.M C'o.1 V ,00v e.(PEAR) OUTPUT VOLTAGE 1000 C.P.S. R'10R cc. RADIO INTEGRATOR OUTPUT 1T JJreR'DIPPERENTIATOROUTPUT 50000 USECONDS R 1lC PERIOD OP e' 1000 USECONDS +100V Figure 28 R -C COUPLING CIRCUIT WITH LONG TIME CONSTANT 100 v et ,-11 INTEGRATOR OUTPUT I E'100v. (PEAR ) I I ófiE[Ñ1TÓR"N eo 100 V +25 V. I 1 10001 -100v eR IATOR DIP OUTPUT +30v PR o AO T OUTPUT Of ATOR o (ec) T Figure 30 RC DIFFERENTIATOR AND INTEGRATOR ACTION ON Figure 29 R -C DIFFERENTIATOR AND INTEGRATOR ACTION ON A SQUARE WAVE A SINE WAVE Sawtooth Wave Input shape of a pure sine wave, they will merely shift the phase of the wave (figure 29). The differentiator output is a sine wave leading the input wave, and the integrator output is a sine wave which lags the input wave. The sum of the two outputs at any instant equals the instantaneous input voltage. If a square wave voltage is impressed on the circuit of figure 30, a square wave voltage output may be obtained across the integrating capacitor if the time constant of the circuit allows the capacitor to become fully charged. In this particular case, the capacitor never fully charges, and as a result the output of the integrator has a smaller amplitude than the input. The differentiator output has a maximum value greater than the input amplitude, since the voltage left on the capacitor from the previous half wave will add to the input voltage. Such a circuit, when used as a differentiator, is often called a peaker. Peaks of twice the input amplitude may be produced. Square Wave Input If a back -to -back saw tooth voltage is applied to an RC circuit having a time constant one sixth the period of the input voltage, the result is shown in figure 31. The capacitor voltage will closely follow the input voltage, if the time constant is short, and the integrator output closely resembles the input. The amplitude is slightly reduced and there is a slight phase lag. Since the voltage across the capacitor is increasing at a constant rate, the charging and discharging current is constant. The output voltage of the differentiator, therefore, is constant during each half of the saw tooth input. voltage waveforms Various other than those represented here may be applied to short RC circuits for the purpose of producing across the resistor an output voltage with an amplitude proportional to the rate of change of the input signal. The shorter the RC time constant is made with respect to the period of the input wave, the more nearly the voltage across Miscellaneous Inputs www.americanradiohistory.com Transformers HANDBOOK e=100 61 INTEGRATOR OUTPUT (ec) v. (PEAR) 1000 C.P.S. 1 DIFFERENTIATOR JOUTPUT - (e0) +100 OUTPUT WAVEFORM OF GENERATOR -100 -ppt{ IO uJ e0 ÌA ERÉITTÓRp I(eR) e0 OUTPUT OF INTEGRATOR (eE) Figure 31 DIFFERENTIATOR AND INTEGRATOR ACTION ON R -C A SAWTOOTH WAVE Figure 32 the capacitor conforms to the input voltage. Thus, the differentiator output becomes of particular importance in very short RC circuits. Differentiator outputs for various types of input waves are shown in figure 32. The application of a square wave input signal to audio equipment, and the observation of the reproduced output signal on an oscilloscope will provide a quick and accurate check of the overall operation of audio equipment. Low -frequency and high- frequency response, as well as transient response can be examined easily. If the amplifier is deficient in low- frequency response, the flat top of the square wave will be canted, as in figure 33. If the high- frequency response is inferior, the rise time of the output wave will be retarded (figure 34). An amplifier with a limited highand low- frequency response will turn the square wave into the approximation of a saw tooth wave (figure 35). Square Wave Test for Audio Equipment Transformers are placed in such inductive coils When two relation to each other that the lines of force 3-4 from one cut across the turns of the other inducing a current, the combination can be called a transformer. The name is derived from the fact that energy is transformed from one winding to another. The inductance in which Dlfferentlator outputs of short r -c circuits for various input voltage waveshapes. The output voltage is proportional to the rote of change of the input voltage. the original flux is produced is called the primary; the inductance which receives the induced current is called the secondary. In a radio receiver power transformer, for example, the coil through which the 110 -volt a.c. passes is the primary, and the coil from which a higher or lower voltage than the a -c line potential is obtained is the secondary. Transformers can have either air or magnetic cores, depending upon the frequencies at which they are to be operated. The reader should thoroughly impress upon his mind the fact that current can be transferred from one circuit to another only if the primary current is changing or alternating. From this it can be seen that a power transformer cannot possibly function as such when the primary is supplied with non -pulsating d.c. A power transformer usually has a magnetic core which consists of laminations of iron, built up into a square or rectangular form, with a center opening or window. The secondary windings may be several in number, each perhaps delivering a different voltage. The secondary voltages will be proportional to the turns ratio and the primary voltage. www.americanradiohistory.com Alternating Current Circuits 62 RADIO THE Figure 33 Amplifier deficient In low frequency response will distort square wave applied to the input circuit, as shown. A 60 -cycle square wave may he used. A: B: C: D: Drop in gain at low frequencies Lending phase shift at low frequencies Logging phase shift at low frequencies Accentuated low frequency gain Types of Transformers Transformers are used in alteroaring- current circuits to transfer power at one voltage and impedance to another circuit at another voltage and impedance. There are three main classifications of transformers: those made for use in power-frequency circuits, those made for audio -frequency applications, and those made for radio frequencies. The Transformation Ratio In a perfect transformer all the magnetic flux lines produced by the primary winding link every turn of the secondary winding. For such a transformer, the ratio of the primary and secondary voltages is exactly the same as the ratio of the number of turns in the two windings: Np across the secondary winding In practice, the transformation ratio of a transformer is somewhat less than the turns ratio, since unity coupling does not exist between the primary and secondary windings. Ampere Turns (NI) The current that flows in the secondary winding as a result of the induced voltage must produce a flux which exactly equals the primary flux. The magnetizing force of a coil is expressed as the product of the number of turns in the coil times the current flowing in it: NpxIp =NsXIs, or Np Ns = Is IP where Ip = primary current Ep Ns Es where Np = number of turns in the primary winding Ns = number of turns in the secondary winding Ep = voltage across the primary winding Figure Es = voltage Is = secondary current It can be seen from this expression that when the voltage is stepped up, the current is stepped down, and vice -versa. Leakage Reactance 34 Since unity coupling does not exist in a practical Figure 35 Output waveshape of amplifier having deficiency in high- frequency response. Tested with 10 -kc. square wave. Output waveshape of amplifier having limited low -frequency and high- frequency response. Tested with 1 -kc. square wave. www.americanradiohistory.com Electric Filters HANDBOOK 63 1 STEP -UP ZL STEP -OOW N INPUT OUTPUT VOLTAGE VOLTAGE Figure 36 IMPEDANCE -MATCHING TRANSFORMER The reflected impedance Zp varies directly In the secondary load IL, and proportion to the square of the primary-to- secondary turns ratio. proportion directly to In transformer, part of the flux passing from the primary circuit to the secondary circuit follows a magnetic circuit acted upon by the primary only. The same is true of the secondary flux. These leakage fluxes cause leakage reactance in the transformer, and tend to cause the transformer to have poor voltage regulation. To reduce such leakage reactance, the primary and secondary windings should be in close proximity to each other. The more expensive transformers have interleaved windings to reduce inherent leakage reactance. Impedance In the ideal transformer, the impedance of the secondary load is reflected back into the primary winding in the following relationship: Transformation Zp = N'Zs , or N = N/Zp/Zs where Zp = reflected primary impedance N = turns ratio of transformer Zs = impedance of secondary load Thus any specific load connected to the secondary terminals of the transformer will be transformed to a different specific value appearing across the primary terminals of the transformer. By the proper choice of turns ratio, any reasonable value of secondary load impedance may be "reflected" into the primary winding of the transformer to produce the desired transformer primary impedance. The phase angle of the primary "reflected" impedance will be the same as the phase angle of the load impedance. A capacitive secondary load will be presented to the transformer source as a capacity, a resistive load will present a resistive "reflection" to the primary source. Thus the primary source "sees" a transformer load entirely dependent upon the secondary load impedance and the turns ratio of the transformer (figure 36). The type of transformer in figure 37, when wound with heavy wire over an iron core, is a common device in primary power circuits for the purpose of increasing or decreasing the line volt- Figure 37 THE AUTO -TRANSFORMER auto- transformer Schematic diagram of an showing the method of connecting it to the line and to the load. When only a small amount of step up or step down Is required, the auto transformer may be much smaller physically thon would be a transformer with o separate Continuously variable winding. secondary auto -transformers (Variar and Powerstat) are widely used commercially. age. In effect, it is merely a continuous winding with taps taken at various points along the winding, the input voltage being applied to the bottom and also to one tap on the winding. If the output is taken from this same tap, the voltage ratio will be 1 -to -1; i.e., the input voltage will be the same as the output voltage. On the other hand, if the output tap is moved down toward the common terminal, there will be a step -down in the turns ratio with a consequent step-down in voltage. The initial setting of the middle input tap is chosen so that the number of turns will have sufficient reactance to keep the no -load primary current at a reasonably low value. Electric Filters 3 -5 There are many applications where it is desirable to pass a d -c component without passing a superimposed a -c component, or to ELEMENTARY FILTER SECTIONS T- NET WONIt L- SECTIONS rs T Pi - NETWOR The Auto Transformer Figure 38 Complex filters may be mode up from these basic www.americanradiohistory.com filter sections. 64 A l t e rn a t n g i C ur r e n LOW -PASS SHUNT -DERIVE t C i rc u i t T H E s R A D I O HIGH-PASS SERIES -DERI ED FILTER FILTER (SERIES -ARM RESONATED (5J.UNT -ARM RESONATE CI 2 2CI 2C1 C2 O < z fQ f2 fq FREQUENCY R. L FREQUENCY LOAD RESISTANCE R. Ci 1 4M x C C2- K C2= MCK LK= f2 = LOAD RESISTANCE CI' M LE L2M= ,/I -()2 CUT -OFF FREQUENCY. Cx= 777_tF- fk =FREQUENCY OF NIGH ATTENUATION LK 14M> -x M- = Cs 7- M-I /ßa`2 ( fI= I coverage book. Filter Operation A filter acts by virtue of its property of offering very high impedance to the undesired frequencies, while offering but little impedance to the desired frequencies. This will also apply to d.c. with a superimposed a -c component, as d.c. can be considered as an alternating current of zero frequency so far as filter discussion goes. Basic Filters Filters are divided into four classes, descriptive of the fre- quency bands which they are designed to transmit: high pass, low pass, band pass and band elimination. Each of these classes of filters is made up of elementary filter sections called L sections which consist of a series element (ZA) and a parallel element (ZR) as 477fIR NIGH ATTENUATION Figure 39 TYPICAL LOW -PASS AND HIGH -PASS FILTERS, ILLUSTRATING DERIVATIONS pass all frequencies above or below a certain frequency while rejecting or attenuating all others, or to pass only a certain band or bands of frequencies while attenuating all others. All of these things can be done by suitable combinations of inductance, capacitance and resistance. However, as whole books have been devoted to nothing but electric filters, it can be appreciated that it is possible only to touch upon them superficially in a general CS Cur-OFF FREQUENCY. P, =FREQUENCY OF SHUNT AND SERIES illustrated in figure 38. A finite number of L sections may be combined into basic filter sections, called T networks or pi networks, also shown in figure 38. Both the T and pi networks may be divided in two to form halfsections. Filter Sections The most common filter section is one in which the two impedances ZA and Zg are so related that their arithmetical product is a constant: ZA x Zg = K2 at all frequencies. This type of filter section is called a constant-K section. A section having a sharper cutoff frequency than a constant -K section, but less attenuation at frequencies far removed from cutoff is the M- derived section, so called because the shunt or series element is resonated with a reactance of the opposite sign. If the complementary reactance is added to the series arm, the section is said to be shunt derived; if added to the shunt arm, series derived. Each impedance of the M- derived section is related to a corresponding impedance in the constant K section by some factor which is a function of the constant m. M, in turn, is a function of the ratio between the cutoff frequency and the frequency of infinite attenuation, and will www.americanradiohistory.com Filter Design HANDBOOK TT- SECTION FILTER DESIGN M=0 CONSTANT K R' LOAD RESISTANCE =CUT-OFF FREQUENCY 2 f.= FREQUENCY OFVERY HIGH ATTENUATION °---/-i C 0 Ln- n R f2 O C2 ltf2 R T 2 f2 RL fICUT-OFFFREQUENCY FREQUENCY OF VERY HIGH ATTENUATION 2L2 LK=4 2L2-- 21_2 4 /Vt I Cl. S1= 0.6 Lz= la o.s R z z ó 0 /¡ ='-( f12'oe ) = 2L2 I 1 2L2 -- }Lt iLt I I .`..1 t I 2L2 0 SAME VALUES ASM=0.6 1--k . 1 fm SAME CURVE AS M < \ c 0.6 z ú i 1 Lt=3.75L1t=t4M xLK C/=Cn L2=LK H IGH PASS Cn- M=0_6 FREQUENCY CI LOAD RESISTANCE o.6 VALUES AS M --f2 < . }SAME SAME CURVE AS M FREQUENCY CK o f fm j j < A 4M Li TiC2 ú T o o z 0 0 6 0 t i C2=o.6Cn=Cn z 0 aC 1 -O Li Ci yI t Lt=0.6Ln=MLK t-M2 Cz-Cn .11-(--f-2-,. 2- f T t------0 C2 T t Cn - R. OTO Lt=Ln LOW PASS M 2 SECTIONSS LFSEC NG -F-YO013y I 65 f, .< r FREQUENCY /f t FREQUENCY Figure 40 Through the use of the curves and equations which accompany the diagrams in the illustration above it is possible to determine the correct values of inductance and capacitance for the usual types of pi- section filters. have some value between zero and one. As the value of m approaches zero, the sharpness of cutoff increases, but the less will be the attenuation at several times cutoff frequency. A value of 0.6 may be used for min most applications. The "notch" frequency is determined by the resonant frequency of the tuned filter element. The amount of attenuation obtained at the "notch" when a derived section is used is determined by the effective Q of the resonant arm (figure 39). Filter Assembly Constant -K sections and derived sections may be cas- caded to obtain the combined characteristics of sharp cutoff and good remote frequency attenuation. Such a filter is known as a composite filter. The amount of attenuation will depend upon the number of filter sections used, and the shape of the transmission curve depends upon the type of filter sections used. All filters have some insertion loss. This attenuation is usually uniform to all frequen- cies within the pass band. The insertion loss varies with the type of filter, the Q of the components and the type of termination employed. Electric wave filters have long been used in some amateur siations in the audio channel to reduce the transmission of unwanted high frequencies and hence to reduce the bandwidth occupied by a radiophone signal. The effectiveness of a properly designed and properly used filter circuit in reducing QRM and sideband splatter should not be underestimated. In recent years, high frequency filters have become commonplace in TVI reduction. High pass type filters are placed before the input stage of television receivers to reject the fundamental signal of low frequency transmitters. Low -pass filters are used in the output circuits of low frequency transmitters to prevent harmonics of the transmitter from being radiated in the television channels. Electric Filter Design www.americanradiohistory.com 66 Alternating Current Circuits The chart of figure 40 gives design data and procedure on the pi- section type of filter. M- derived sections with an M of 0.6 will be found to be most satisfactory as the input section (or half- section) of the usual filter since the input impedance of such a section is most constant over the pass band of the filter section. Simple filters may use either L, T, or n sections. Since the rr section is the more commonly used type figure 40 gives design data and characteristics for this type of filter. A PUSH -PULL 250 -TH AMPLIFIER WITH TVI SHIELD REMOVED filters in power leads and antenna circuit reduces radiation of TVI- producing harmonics of typical push -pull amplifier. Shielded enclosure completes harmonic reduction measures. Use of harmonic www.americanradiohistory.com CHAPTE:R FOUR Vacuum Tube Principles electron tubes the cathode energy is applied in the form of heat; electron emission from a heated cathode is called tbermionic emission. In another common type of electron tube, the photoelectric cell, energy in the form of light is applied to the cathode to cause photoelectric emission. In the previous chapters we have seen the manner in which an electric current flows through a metallic conductor as a result of an electron drift. This drift, which takes place when there is a difference in potential between the ends of the metallic conductor, is in addition to the normal random electron motion between the molecules of the conductor. The electron may be considered as a minute negatively charged particle, having a mass of 9 x 10 -7° gram, and a charge of 1.59 x 10 -19 coulomb. Electrons are always identical, regardless of the source from which they are Thermionic Emission 4-1 of electrons from the cathode of a thermionic electron tube takes place when the cathode of the tube is heated to a temperature sufficiently high that the free electrons in the emitter have sufficient velocity to overcome the restraining forces at the surface of the material. These surface forces vary greatly with different materials. Hence different types of cathodes must be raised to different temperatures to obtain adequate quantities of electron emission. The several types of emitters found in common types of transmitting and receiving tubes will be described in the following paragraphs. Electron Emission obtained. An electric current can be caused to flow through other media than a metallic conductor. One such medium is an ionized solution, such as the sulfuric acid electrolyte in a storage battery. This type of current flow is called electrolytic conduction. Further, it was shown at about the turn of the century that an electric current can be carried by a stream of free electrons in an evacuated chamber. The flow of a current in such a manner is said to take place by electronic conduction. The study of electron tubes (also called vacuum tubes, or valves) is actually the study of the control and use of electronic currents within an evacuated or partially evacuated chamber. Since the current flow in an electron tube takes place in an evacuated chamber, there must be located within the enclosure both a source of electrons and a collector for the electrons which have been emitted. The electron source is called the cathode, and the electron collector is usually called the anode. Some external source of energy must be applied to the cathode in order to impart sufficient velocity to the electrons within the cathode material to enable them to overcome the surface forces and thus escape into the surrounding medium. In the usual types of Emission Cathode Types The emitters or cathodes as used in present -day thermionic electron tubes may be classified into two groups: the directly- heated or filament type and the indirectly -heated or heater - cathode type. Directly- heated emitters may be further subdivided into three important groups, all of which are commonly used in modern vacuum tubes. These classifications are: the puretungsten filament, the thoriated- tungsten filament, and the oxide-coated filament. The Pure Tung- sten Filament 67 www.americanradiohistory.com Pure tungsten wire was used as the filament in nearly all the earlier transmitting and 68 Vacuum Tube THE Principles RADIO Figure ELECTRON TUBE TYPES The new General E l e c t r i c ceramic triode (68Y4) is shown alongside a conventional miniature tube (6265) and an octal -based receiving tube (25L6). The ceramic tube is designed for rugged service and features extremely low lead inductance. 1 receiving tubes. However, the thermionic efficiency of tungsten wire as an emitter (the number of milliamperes emission per watt of filament heating power) is quite low, the filaments become fragile after use, their life is rather short, and they are susceptible to burnout at any time. Pure tungsten filaments must be run at bright white heat (about 2500° Kelvin). For these reasons, tungsten filaments have been replaced in all applications where another type of filament could be used. They are, however, still universally employed in large water-cooled tubes and in certain large, high -power air- cooled triodes where another filament type would be unsuitable. Tungsten filaments are the most satisfactory for high power, high -voltage tubes where the emitter is subjected to positive ion bombardment caused by the residual gas content of the tubes. Tungsten is not adversely affected by such bombardment. the course of experiments made upon tungsten emitters, it was found that filaments made from tungsten having a small amount of thoria (thorium oxide) as an impurity had much greater emission than those made from the pure metal. Subsequent development has resulted in the highly efficient carburized thoriated- tungsten filament as used in virtually all medium -power transmitting tubes today. Thoriated-tungsten emitters consist of a tungsten wire containing from 1% to 2% thoria. The activation process varies between different manufacturers of vacuum tubes, but it is essentially as follows: (1) the tube is evacuated; (2) the filament is burned for a short period at about 2800° Kelvin to clean the surface and reduce some of the thoria within the filament to metallic thorium; (3) The ThoriatedTungsten Filament In the filament is burned for a longer period at about 2100° Kelvin to form a layer of thorium on the surface of the tungsten; (4) the temperature is reduced to about 1600° Kelvin and some pure hydrocarbon gas is admitted to form a layer of tungsten carbide on the surface of the tungsten. This layer of tungsten carbide reduces the rate of thorium evaporation from the surface at the normal operating temperature of the filament and thus increases the operating life of the vacuum tube. Thorium evaporation from the surface is a natural consequence of the operation of the thoriatedtungsten filament. The carburized layer on the tungsten wire plays another role in acting as a reducing agent to produce new thorium from the thoria to replace that lost by evaporation. This new thorium continually diffuses to the surface during the normal operation of the filament. The last process, (5), in the activation of a thoriated tungsten filament consists of re- evacuating the envelope and then burning or ageing the new filament for a considerable period of time at the normal operating temperature of approximately 1900°K. One thing to remember about any type of filament, particularly the thoriated type, is that the emitter deteriorates practically as fast when "standing by" (no plate current) as it does with any normal amount of emission load. Also, a thoriated filament may be either temporarily or permanently damaged by a heavy overload which may strip the surface layer of thorium from the filament. Thoriated- tungsten filaments (and only thoriatedtungsten filaments) which have lost emission as a result of insufficient filament voltage, a severe temporary overload, a less severe extended overload, or even normal operation Reactivating Thoriated- Tungsten Filaments www.americanradiohistory.com Types of Emitters HANDBOOK Figure 69 2 V -H -F and U -H -F TUBE TYPES The tube to the left In this photograph is a 955 "acorn" triode. The 6F4 acorn triode is very similar in appearance to the 955 but has two leads brought out each for the grid and for the plate connection. The second tubs Is a 446A "lighthouse" triode. The 2C40, 2C43, and 2C44 ore more recent examples of the same type tube and are tially the same in external appearance. The third tube from the left is o 2C39 "oilcan" tube. This tube type is essentially the inverse of the lighthouse variety since the cathode and heater connections come out the small end and the plats is the large finned radiator on the large end. The use of the finned plate radiator makes the oilcan tube capable of approximately 10 times as much plate dissipation as the lighthouse type. The tube to the right is the 4X 150A beam tetrads. This tube, a comparatively recent release, is capable of somewhat greater power output than any of the other tube types shown, and is rated for full output at 500 Mc. and of reduced output at frequencies greater than 1000 Mc. may quite frequently be reactivated to their original characteristics by a process similar to that of the original activation. However, only filaments which have not approached too close to the end of their useful life may be successfully reactivated. The actual process of reactivation is relatively simple. The tube which has gone "flat" is placed in a socket to which only the two filament wires have been connected. The filament is then "flashed" for about 20 to 40 seconds at about 1% times normal rated voltage. The filament will become extremely bright during this time and, if there is still some thoria left in the tungsten and if the tube did not originally fail as a result of an air leak, some of this thoria will be reduced to metallic thorium. The filament is then burned at 15 to 25 per cent overvoltage for from 30 minutes to 3 to 4 hours to bring this new thorium to the surface. The tube should then be tested to see if it shows signs of renewed life. If it does, but is still weak, the burning process should be continued at about 10 to 15 per cent overvoltage for a few more hours. This should bring it back almost to normal. If the tube checks still very low after the first attempt at reactivation, the complete process can be repeated as a last effort. The Oxide. Coated Filament The most efficient of all modern filaments is the oxide-coated type which con- sists of a mixture of barium and strontium oxides coated upon a nickel alloy wire or strip. This type of filament operates at a dull red to orange -red temperature (1050' to 1170° K) at which temperature it will emit large quantities of electrons. The oxide -coated filament is somewhat more efficient than the thoriated- tungsten type in small sizes and it is considerably less expensive to manufacture. For this reason all receiving tubes and quite a number of the low -powered transmitting tubes use the oxide - coated filament. Another advantage of the oxide -coated emitter is its the average tube can be extremely long life - expected to run from 3000 to 5000 hours, and when loaded very lightly, tubes of this type have been known to give 50,000 hours of life before their characteristics changed to any great extent. Oxide filaments are unsatisfactory for use at high continuous plate voltages because: (1) their activity is seriously impaired by the high temperature necessary to de -gas the high voltage tubes and, (2) the positive ion bombardment which takes place even in the best evacuated high -voltage tube causes destruction of the oxide layer on the surface of the filament. Oxide -coated emitters have been found capable of emitting an enormously large current pulse with a high applied voltage for a very short period of time without damage. This characteristic has proved to be of great value www.americanradiohistory.com 70 Vacuum Tube Figure 3 CUTAWAY DRAWING OF A THE Principles 6C4 TRIODE RADIO age from 2 to 117 volts, although 6.3 is the most common value. The heater is operated at quite a high temperature so that the cathode itself usually may be brought to operating temperature in a matter of 15 to 30 seconds. Heat -coupling between the heater and the cathode is mainly by radiation, although there is some thermal conduction through the insulating coating on the heater wire, as this coating is also in contact with the cathode thimble. Indirectly heated cathodes are employed in all a-c operated tubes which are designed to operate at a low level either for r-f or a -f use. However, some receiver power tubes use heater cathodes (6L6, 6V6, 6F6, and 6K6 -GT) as do some of the low-power transmitter tubes (802, 807, 815, 3E29, 2E26, 5763, etc.). Heater cathodes are employed almost exclusively when a number of tubes are to be operated in series as in an a.c. -d.c. receiver. A heater cathode is often called a uni- potential cathode because there is no voltage drop along its length as there is in the directly- heated or filament cathode. in radar work. For example, the relatively small cathode in a microwave magnetron may be called upon to deliver 25 to 50 amperes at an applied voltage of perhaps 25,000 volts for a period in the order of one microsecond. After this large current pulse has been passed, plate voltage normally will be removed for 1000 microseconds or more so that the cathode surface may be restored in time for the next pulse of current. If the cathode were to be subjected to a continuous current drain of this magnitude, it would be destroyed in an exceedingly short period of time. The activation of oxide- coated filaments also varies with tube manufacturers but consists essentially in heating the wire which has been coated with a mixture of barium and strontium carbonates to a temperature of about 1500° Kelvin for a time and then applying a potential of 100 to 200 volts through a protective resistor to limit the emission current. This process reduces the carbonates to oxides thermally, cleans the filament surface of foreign materials, and activates the cathode special bombardment cathode is employed in many of the new high powered television transmitting klystrons(Eimac 3K 20,000 LA). The cathode takes the form of a tantalum diode, heated to operating temperature by the bombardment of electrons from a directly heated filament. The cathode operates at a positive potential of 2000 volts with respect to the filament, and a d-c bombardment current of 0.66 amperes flows between filament and cathode. The filament is designed to operate under space -charge limited conditions. Cathode temperature is varied by changing the bombardment potential between the filament and the cathode. The heater type cathode was developed as a result of the requirement for a type of emitter which could be operated from alternating current and yet would not introduce a-c ripple modulation even when used in low -level stages. It consists essentially of a small nickel -alloy cylinder with a coating of strontium and barium oxides on its surface similar to the coating used on the oxide- coated filament. Inside the cylinder is an insulated heater element consisting usually of a double spiral of tungsten wire. The heater may operate on any volt- The emission of electrons from a heated cathode is quite similar to the evaporation of molecules from the surface of a liquid. The molecules which leave the surface are those having sufficient kinetic (heat) energy to overcome the forces at the surface of the liquid. As the temperature of the liquid is raised, the average velocity of the molecules is increased, and a greater number of molecules will acquire sufficient energy to be evaporated. The evaporation of electrons from the surface of a thermionic emitter is similarly a function of average electron velocity, and hence is a function of the temperature of the emitter. Electron emission per unit area of emitting surface is a function of the temperature T in degrees Kelvin, the work function of the emitting surface b (which is a measure of the surface. Reactivation of oxide- coated filaments is not possible since there is always more than sufficient reduction of the oxides and diffusion of the metals to the surface of the filament to meet the emission needs of the cathode. The Heater Cathode The Bombardment Cathode The Emission Equation www.americanradiohistory.com A Thermionic Emission HANDBOOK 71 600 TYPE 6CB6 6W4-GT Er' 63 VOLTS / PLATE 600 35LP?RS314R w ÌSCREEN W 4910 __-, frT. ¢ 400 çArHOOE HEATER HEATER L - - - -- i , F ww 200 á Figure 4 CUT -AWAY DRAWING OF A 6CB6 PENTODE surface forces of the material and hence of the energy required of the electron before it may escape), and of the constant A which also varies with the emitting surface. The relationship between emission current in amperes per square centimeter, 1, and the above quantities can be expressed as: = AT'c"b'T Secondary The bombarding of most metals and a few insulators by electrons Emission will result in the emission of other 1 electrons by a process called secondary emis- The secondary electrons are literally knocked from the surface layers of the bombarded material by the primary electrons which strike the material. The number of secondary electrons emitted per primary electron varies from a very small percentage to as high as 5 to 10 secondary electrons per primary. The phenomena of secondary emission is undesirable for most thermionic electron tubes. However, the process is used to advantage in certain types of electron tubes such as the image orthicon (TV camera tube) and the electron -multiplier type of photo -electric cell. In types of electron tubes which make use of secondary emission, such as the type 931 photo cell, the secondary- electron -emitting surfaces are specially treated to provide a high ratio of secondary to primary electrons. Thus a high degree of current amplification in the electron -multiplier section of the tube is sion. obtained. As a cathode is heated so that it begins to emit, those electrons which have been discharged into the surrounding space form a negatively charged cloud in the immediate vicinity of the cathode. This cloud of electrons around the cathode is called the space charge. The electrons comprising the charge are conThe Space Charge Effect tinuously changing, since those electrons making up the original charge fall back into 20 10 30 ao so D.C. PLATE VOLTS Figure 5 AVERAGE PLATE CHARACTERISTICS OF A POWER DIODE the cathode and are replaced by others emitted by it. 4 -2 The Diode If a cathode capable of being heated either indirectly or directly is placed in an evacuated envelope along with a plate, such a two element vacuum tube is called a diode. The diode is the simplest of all vacuum tubes and is the fundamental type from which all the others are derived. the cathode within a diode is heated, it will be found that a few of the electrons leaving the cathode will leave with sufficient velocity to reach the plate. If the plate is electrically connected back to the cathode, the electrons which have had sufficient veloc= ity to arrive at the plate will flow back to the cathode through the external circuit. This small amount of initial plate current is an effect found in all two -element vacuum tubes. If a battery or other source of d -c voltage is placed in the external circuit between the plate and cathode so that it places a positive potential on the plate, the flow of current from the cathode to plate will be increased. This is due to the strong attraction offered by the positively charged plate for any negatively charged particles (figure 5). Characteristics of the Diode When At moderate values of plate voltage the current flow from cathode to anode is limited by the space charge of electrons around the cathode. Increased values Space- Charge Limited Current www.americanradiohistory.com Vacuum 72 Principles Tube THE RADIO DE COATED ED TUNGSTEN TUNGSTEN FILAMENT POINT OF MAXIMUM SPACE CHARGE -LIMITED EMISSION Figure ACTION PLATE VOLTAGE Figure + 6 MAXIMUM SPACE -CHARGE -LIMITED EMISSION FOR DIFFERENT TYPES OF EMITTERS of plate voltage will tend to neutralize a greater portion of the cathode space charge and hence will cause a greater current to flow. Under these conditions, with plate current limited by the cathode space charge, the plate current is not linear with plate voltage. In fact it may be stated in general that the plate current flow in electron tubes does not obey Ohm's Law. Rather, plate current increases as the three -halves power of the plate voltage. The relationship between plate voltage, E, and plate current, 1, can be expressed as: / =K F3!2 where K is a constant determined by the geometry of the element structure within the electron tube. As plate voltage is raised to the potential where the cathode space charge is neutralized, all the electrons that the cathode is capable of emitting are being attracted to the plate. The electron tube is said then to have reached saturation plate current. Further increase in plate voltage will cause only a relatively small increase in plate current. The initial point of plate current saturation is sometimes called the point of Maximum Space Charge- Limited Emission (MSCLE). The degree of flattening in the plate -voltage plate- current curve after the MSCLE point will vary with different types of cathodes. This effect is shown in figure 6. The flattening is quite sharp with a pure tungsten emitter. With thoriated tungsten the flattening is smoothed somewhat, while with an oxide- coated cathode the flattening is quite gradual. The gradual saturation in emission with an oxide- coated emitter is generally considered to result from Plate Current Saturation 7 OF THE GRID IN A TRIODE (A) shows the triode tube with cutoff bias on the grid. Note that all the electrons emitted by the cathode remain inside the grid mesh. (B) shows the same tube with an intermediate value of bias on the grid. Note the medium value of plate current and the fact that there is a reserve of electrons remaining within the grid mesh. (C) shows the operation with a relatively small amount of bias which with certain tube types will allow substantially all the electrons emitted by the cathode to reach the plate. Emission is said to be saturated in this case. In a majority of tube types a high value of positive grid voltage is required before plate - current saturation takes place. a lowering of the surface work function by the field at the cathode resulting from the plate potential. Electron Energy Dissipation The current flowing in the plate- cathode space of a conducting electron tube represents the energy required to accelerate electrons from the zero potential of the cathode space charge to the potential of the anode. Then, when these accelerated electrons strike the anode, the energy associated with their velocity is immediately released to the anode structure. In normal electron tubes this energy release appears as heating of the plate or anode structure. 4-3 The Triode If an element consisting of a mesh or spiral of wire is inserted concentric with the plate and between the plate and the cathode, such an element will be able to control by electrostatic action the cathode -to -plate current of the tube. The new element is called a grid, and a vacuum tube containing a cathode, grid, and plate is commonly called a triode. Action of If this new element through which the electrons must pass in their course from cathode to plate is made negative with respect to the cathode, the negathe Grid www.americanradiohistory.com HANDBOOK mom TYPE N 4 MI 6J5 Er=6.5VOLTS A litiliI MM1I 1111111 italar ! A /!!. .111! ..01 ; ! /I AMÍGlI o 100 test. pp. 200 300 IM 400 Current Flow in a Triode 500 PLATE VOLTS (EP) Figure 8 NEGATIVE -GRID CHARACTERISTICS(Ip VS. Ep CURVES) OF A TYPICAL TRIODE Average plate characteristics of this type are most commonly used in determining the Class A operating characteristics of a triode amplifier stage. cive charge on this grid will effectively repel the negatively charged electrons (like charges repel; unlike charges attract) back into the space charge surrounding the cathode Hence, the number of electrons which are able to pass through the grid mesh and reach the plate will be reduced, and the plate current will be reduced accordingly. If the charge on the grid is made sufficiently negative, all the electrons leaving the cathode will be repelled back to it and the plate current will be reduced to zero. Any d-c voltage placed upon a grid is called a bias (especially so when speaking of a control grid). The smallest negative voltage which will cause cutoff of plate current at a particular plate voltage is called the value of cutoff bias (figure 7). Amplification The amount of plate current in a Factor triode is a result of the net field at the cathode from interaction between the field caused by the grid bias and that caused by the plate voltage. Hence, both grid bias and plate voltage affect the plate current. In all normal tubes a small change in grid bias has a considerably greater effect than a similar change in plate voltage. The ratio between the change in grid bias and the change in plate current which will cause the same small change in plate current is called the amplification /actor or of the electron tube. Expressed as an equation: AE - AE, 73 with i, constant (A represents a small increment). The µ can be determined experimentally by making a small change in grid bias, thus slightly changing the plate current. The plate current is then returned to the original value by making a change in the plate voltage. The ratio of the change in plate voltage to the change in grid voltage is the µ of the tube under the operating conditions chosen for the 16 s! Characteristics Triode In a diode it was shown that the electrostatic field at the cathode was proportional to the plate potential, Ep, and that the total cathode current was proportional to the three halves power of the plate voltage. Similiarly, in a triode it can be shown that the field at the cathode space charge is proportional to the equivalent voltage (Eg + Ep /ft), where the amplification factor, µ, actually represents the relative effectiveness of grid potential and plate potential in producing a field at the cathode. It would then be expected that the cathode current in a triode would be proportional to the three -halves power of (E5 + Ep /µ). The cathode current of a triode can be represented with fair accuracy by the expression: Cathode current = K (E5 E 3/2 +-=-) ) where K is a constant determined by element geometry within the triode. Plate Resistance The plate resistance of a vacuum tube is the ratio of a change in plate voltage to the change in plate current which the change in plate voltage produces. To be accurate, the changes should be very small with respect to the operating values. Expressed as an equation: Rp = AE P p E,= constant, A = small increment The plate resistance can also be determined by the experiment mentioned above. By noting the change in plate current as it occurs when the plate voltage is changed (grid voltage held constant), and by dividing the latter by the former, the plate resistance can be determined. Plate resistance is expressed in Ohms. The mutual conductance, also referred to as trans conductance, is the ratio of a change in the plate current to the change in grid voltage which brought about the plate current change, the plate voltage being held constant. Expressed as an equation: Transconductance www.americanradiohistory.com Tube Vacuum 74 Principles RADIO THE 430 400 330 300 2 110 _ W 6 6 seo W 130 C 100 -20 o -0 -e -10 10 10 b /0 40 10 GRID VOLTAGE (E9) 70 e0 10 100 characteristics of this type are most commonly used in determining the pulse -signal operating characteristics of a triode amplifier stage. Note the large emission capability of the oxidecoated heater cathode in tubes of the type of the 6J5. 9 Al constant, small increment The transconductance is also numerically equal to the amplification factor divided by the plate resistance. Gm = E = A = AEg Transconductance is most commonly expressed in microreciprocal -ohms or micro mhos. However, since transconductance expresses change in plate current as a function of a change in grid voltage, a tube is often said to have a transconductance of so many milliamperes- per-volt. If the transconductance in milliamperes -per-volt is multiplied by 1000 it will then be expressed in micromhos. Thus the transconductance of a 6A3 could be called either 5.25 ma.!volt or 5250 micromhos. 0 1-5 (Es) Figure 10 This type of graphical for Class C amplifier operating characteristic is a straight line when Plate 1 -10 -5 GRID VOLTS CONSTANT CURRENT (Ep VS. E9) CHARACTERISTICS OF A TYPICAL TRIODE TUBE Figure 9 POSITIVE -GRID CHARACTERISTICS (Ip VS. E9) OF A TYPICAL TRIODE Gm = -11 representation is used calculations since the of a Class C amplifier drawn upon a constant current- graph. passing through the plate circuit of the tube for various values of plate -load resistance and plate - supply voltage. Figure 11 illustrates a triode tube with a resistive plate load, and a supply voltage of 300 volts. The voltage at the plate of the tube (ep) may be expressed as: ep = Ep -(i x RL) where Ep is the plate supply voltage, ip is the plate current, and RL is the load resistance in ohms. Assuming various values of ip flowing in the circuit, controlled by the internal resistance of the tube, (a function of the grid bias) values of plate voltage may be plotted as shown for each value of plate current (ir). The line connecting these points is called the load line for the particular value of plate -load resistance used. The slope of the load line is equal to the ratio of the lengths of the vertical and horizontal projections of any segment of the load line. For this example it is: The operating character istics of a triode tube may be summarized in three sets of curves: The Ip vs. Ep curve (figure 8), the Ip vs. Eg curve (figure 9) and the Ep vs. E curve (figure 10). The plate resistance (Rj of the tube may be observed from the Ip vs. Ep curve, the transconductance (Gm) may be observed from the Ip vs. Eg curve, and the amplification factor (pt) may be determined from the Ep vs. Eg curve. The slope of the load line is equal to -1/11L. At point A on the load line, the voltage across the tube is zero. This would be true for a perfect tube with zero internal voltage drop, or if the tube is short -circuited from cathode to plate. Point B on the load line load line is a graphical representation of the voltage on the plate of a vacuum tube, and the current corresponds to the cutoff point of the tube, where no plate current is flowing. The operating range of the tube lies between these two extremes. For additional information re- Characteristic Curves of a Triode Tube The Load Line A Slope = - www.americanradiohistory.com .01 100 - .02 200 - .0001 - 1 10,000 Triode Load Line HANDBOOK IP(YA)i En S 10 IS 20 25 30 EP=300v I 300 250 200 I 75 RL=en SO 100 50 0 Figure 12 TRIODE TUBE CONNECTED FOR DETERMINATION OF PLATE CIRCUIT LOAD LINE, AND OPERATING PARAMETERS OF THE CIRCUIT 300 EP Figure 11 voltage drop across the plate load resistor, RL. The plate voltage on the tube is therefore 300 volts. If, on the other hand, the tube is considered to be a short circuit, maximum possible plate current flows and the full 300 volt drop appears across RL. The plate voltage is zero, and the plate current is 300/8,000, or 37.5 milliamperes. These two extreme conditions define the load line on the I, vs. Ep characteristic curve, figure 13. For this application the grid of the tube is returned to a steady biasing voltage of -4 volts. The steady or quiescent operation of the tube is determined by the intersection of the load line with the -4 volt curve at point Q. By projection from point Q through the plate The static load line for a typical triode tube with a plate load resistance of 10,000 ohms. garding dynamic load lines, the reader is referred to the Radiotron Designer's Handbook, 4th edition, distributed by Radio Corporation of America. Application of Tube Characteristics As an example of the ap- plication of tube characteristics, the constants of the triode amplifier circuit shown in figure 12 volts, and the plate load is 8,000 ohms. If the tube is considered to be an open circuit no plate current will flow, and there is no may be considered. The plate supply is 40 37.5 3s 0J. 30 , --i , .. LOGO LINO 4000n iAJ 1 2 Figure = EG fL INSTANTANEOUS SWING 13 i 2 2 APPLICATION OF Ip VS. Ep CHARACTERISTICS OF VACUUM TUBE IP11145-410.2 dFt ,t' -- -- 14N. o leo x 2ee s 300 400 PLATE VOLTS (E.) N www.americanradiohistory.com VOLT Pt.AT[ SWING ` Vacuum 76 EG -4r Tube \ Principles . D.C. THE RADIO BIAS LEVEL (EC) TFigure 15 SCHEMATIC REPRESENTATION + 18.23 OF STEADY STATE // PLATE CURRENT\i)1 INTERELECTRODE CAPACITANCE +is comes apparent. A voltage variation of 8 volts (peak -to -peak) on the grid produces a variation of 84 volts at the plate. T- Polarity Inversion STEADY STATE (EP) PLATE VOLTAGE EP Figure 14 POLARITY REVERSAL BETWEEN GRID AND PLATE VOLTAGES current axis it is found that the value of plate current with no signal applied to the grid is 12.75 milliamperes. By projection from point Q through the plate voltage axis it is found that the quiescent plate voltage is 198 volts. This leaves a drop of 102 volts across RL which is borne out by the relation 0.01275 x 8,000 = 102 volts. An alternating voltage of 4 volts maximum swing about the normal bias value of -4 volts is applied now to the grid of the triode amplifier. This signal swings the grid in a positive direction to 0 volts, and in a negative direction to -8 volts, and establishes the operating region of the tube along the load line between points A and B. Thus the maxima and minima of the plate voltage and plate current are established. By projection from points A and B through the plate current axis the maximum instantaneous plate current is found to be 18.25 milliamperes and the minimum is 7.5 milliamperes. By projections from points A and B through the plate voltage axis the minimum instantaneous plate voltage swing is found to be 154 volts and the maximum is 240 volts. By this graphical application of the IP vs. Ep characteristic of the 6SN7 triode the operation of the circuit illustrated in figure 12 be- signal voltage applied to the grid has its maximum positive instantaneous value the plate current is also maximum. Reference to figure 12 shows that this maximum plate current flows through the plate load resistor RL, producing a maximum voltage drop across it. The lower end of RL is connected to the plate supply, and is therefore held at a constant potential of 300 volts. With maximum voltage drop across the load resistor, the upper end of RL is at a minimum instantaneous voltage. The plate of the tube is connected to this end of RL and is therefore at the same minimum instantaneous potential. This polarity reversal between instantaneous grid and plate voltages is further clarified by a consideration of Kirchhoff's law as it applies to series resistance. The sum of the IR drops around the plate circuit must at all times equal the supply voltage of 300 volts. Thus when the instantaneous voltage drop across RL is maximum, the voltage drop across the tube is minimum, and their sum must equal 300 volts. The variations of grid voltage,plate current and plate voltage about their steady state values is illustrated in figure 14. When the Capacitance always exists between any two pieces of metal separated by a dielectric. The exact amount of capacitance depends upon the size of the metal pieces, the dielectric between them, and the type of dielectric. The electrodes of a vacuum tube have a similar characteristic known as the interelectrode capacitance, illustrated in figure 15. These direct capacities in a triode are: grid-tocathode capacitance, grid -to-plate capacitance, and plate -to- cathode capacitance. The interelectrode capacitance, though very small, has a coupling effect, and often can cause unbalance in a particular circuit. At very high Interelectrode Capacitance www.americanradiohistory.com Tetrodes HANDBOOK Pentodes and TYPE 24-A 77 G=-3 esc =so v. u cr u 6 TYPE 6SK7 esc= too v. esu =ov. ` O. 4 4 J 200 300 S00 100 VOLTS (Eel Figure TYPICAL 16 200 300 VOLTS (E) Figure TETRODE CHARACTERISTIC CURVES Ip VS. Ep frequencies (v-h -f), interelectrode capacities become very objectionable and prevent the use of conventional tubes at these frequencies. Special v -h -f tubes must be used which are characterized by very small electrodes and close internal spacing of the elements of the tube. 400 500 17 TYPICAL IP VS. EP PENTODE CHARACTERISTIC CURVES the electrons pass through it and on to the plate. Due also to the screen, the plate current is largely independent of plate voltage, thus making for high amplification. When the screen voltage is held at a constant value, it is possible to make large changes in plate voltage without appreciably affecting the plate current, (figure 16). 4 -4 Tetrode or Screen Grid Tubes Many desirable characteristics can be obtained in a vacuum tube by the use of more than one grid. The most common multi -element tube is the tetrode (four electrodes). Other tubes containing as many as eight electrodes are available for special applications. The quest for a simple and easily usable method of eliminating the effects of the grid-to -plate capacitance of the triode led to the development of the screen grid tube or tetrode. When another grid is added between the grid and plate of a vacuum tube the tube is called a tetrode, and because the new grid is called a screen, as a result of its screening or shielding action, the tube is often called a screen -grid tube. The interposed screen grid acts as an electrostatic shield between the grid and plate, with the consequence that the grid-to -plate capacitance is reduced. Although the screen grid is maintained at a positive voltage with respect to the cathode of the tube, it is maintained at ground potential with respect to r.f. by means of a by-pass capacitor of very low reactance at the frequency of operation. In addition to the shielding effect, the screen grid serves another very useful purpose. Since the screen is maintained at a positive potential, it serves to increase or accelerate the flow of electrons to the plate. There being large openings in the screen mesh, most of The Tetrode When the electrons from the cathode approach the plate with sufficient velocity, they dislodge electrons upon striking the plate. This effect of bombarding the plate with high velocity electrons, with the consequent dislodgement of other electrons from the plate, gives rise to the condition of secondary emission which has been discussed in a previous paragraph. This effect can cause no particular difficulty in a triode because the secondary electrons so emitted are eventually attracted back to the plate. In the screen -grid tube, however, the screen is close to the plate and is maintained at a positive potential. Thus, the screen will attract these electrons which have been knocked from the plate, particularly when the plate voltage falls to a lower value than the screen voltage, with the result that the plate current is lowered and the amplification is decreased. In the application of tetrodes, it is necessary to operate the plate at a high voltage in relation to the screen in order to overcome these effects of secondary emission. The undesirable effects of secondary emission from the plate can be greatly reduced if yet another element is added between the screen and plate. This additional element Is called a suppressor, and tubes in which it is used are called pentodes. The suppressor grid is sometimes connected to the cathode within the tube; sometimes it is brought out to a connecting pin on the tube base, but in any case it is established negaThe Pentode www.americanradiohistory.com Vacuum 78 Tube Principles THE RADIO GRiD - C HODE .L- Cÿ REMOTE CUT -OFF GRID SHARP CUT -OFF GRID - Figure 18 REMOTE CUTOFF GRID STRUCTURE tive with respect to the minimum plate voltage. The secondary electrons that would travel to the screen if there were no suppressor are diverted back to the plate. The plate current is, therefore, not reduced and the amplification possibilities are increased (figure 17). Pentodes for audio applications are designed so that the suppressor increases the limits to which the plate voltage may swing; therefore the consequent power output and gain can be very great. Pentodes for radio frequency service function in such a manner that the suppressor allows high voltage gain, at the same time permitting fairly high gain at low plate voltage. This holds true even if the plate voltage is the same or slightly lower than the screen voltage. Remote cutoff tubes (variable are screen grid tubes in which the control grid structure has been physically modified so as to cause the plate current of the tube to drop off gradually, rather than to have a well defined cutoff point (figure 18). A non -uniform control grid structure is used, so that the amplification factor is different for different parts of the Remote Cutoff mu) Tubes control grid. Remote cutoff tubes are used in circuits where it is desired to control the amplification by varying the control grid bias. The characteristic curve of an ordinary screen grid tube has considerable curvature near the plate current cutoff point, while the curve of a remote cutoff tube is much more linear (figure 19). The remote cutoff tube minimizes cross- talk interference that would otherwise be produced. Examples of remote cutoff tubes are: 6BD6, 6K7, 6SG7 and 6SK7. beam power tube makes use of another method for suppressing secondary emission. In this tube there are four electrodes: a cathode, a grid, a screen, and a plate, so spaced and placed that secondary emission from the plate is suppressed without actual power loss. Because Beam Power Tubes A GRID VOLTS Figure 19 A REMOTE CUTOFF GRID STRUCTURE ACTION OF of the manner in which the electrodes are spaced, the electrons which travel to the plate are slowed down when the plate voltage is low, almost to zero velocity in a certain region between screen and plate. For this reason the electrons form a stationary cloud, or space charge. The effect of this space charge is to repel secondary electrons emitted from the plate and thus cause them to return to the plate. In this way, secondary emission is suppressed. Another feature of the beam power tube is the low current drawn by the screen. The screen and the grid are spiral wires wound so that each turn in the screen is shaded from the cathode by a grid turn. This alignment of the screen and the grid causes the electrons to travel in sheets between the turns of the screen so that very few of them strike the screen itself. This formation of the electron stream into sheets or beams increases the charge density in the screen -plate region and assists in the creation of the space charge in this region. Because of the effective suppressor action provided by the space charge, and because of the low current drawn by the screen, the beam power tube has the advantages of high power output, high power -sensitivity, and high efficiency. The 6L6 is such a beam power tube, designed for use in the power amplifier stages of receivers and spec -h amplifiers or modulators. Larger tubes employing the beam -power principle are being made by various manufacturers for use in the radio -frequency stages of transmitters. These tubes feature extremely high power- sensitivity (a very small amount of driving power is required for a large output), good plate efficiency, and low grid -toplate capacitance. Examples of these tubes are 813, 4 -250A, 4X150A, etc. The grid- screen mu factor (Ass) is analogous to the amplification factor in a triode, except that the screen of a pentode or tetrode is subGrid -Screen Mu Factor www.americanradiohistory.com HANDBOOK Mixer and Converter stituted for the plate of a triode. µ5g denotes the ratio of a change in grid voltage to a change in screen voltage, each of which will produce the same change in screen current. Expressed as an equation: AEss flag = AEs Ise = constant, A = small increment The grid- screen mu factor is important in determining the operating bias of a tetrode or pentode tube. The relationship between control -grid potential and screen potential determines the plate current of the tube as well as the screen current since the plate current is essentially independent of the plate voltage in tubes of this type. In other words, when the tube is operated at cutoff bias as determined by the screen voltage and the grid screen mu factor (determined in the same way as with a triode, by dividing the operating voltage by the mu factor) the plate current will be substantially at cutoff, as will be the screen current. The grid- screen mu factor is numerically equal to the amplification factor of the same tetrode or pentode tube when it is triode connected. The following equation is the expression for total cathode cur rent in a triode tube. The expression for the total cathode current of a tetrode and a pentode tube is the same, except that the screen -grid voltage and the grid- screen it-factor are used in place of the plate voltage and it of the triode. Current Flow in Tetrodes and Pentodes Cathode current = K / 1 E Es + 3/2 $g ) Ilse Cathode current, of course, is the sum of the screen and plate current, plus control grid current in the event that the control grid is positive with respect to the cathode. It will be noted that total cathode current is independent of plate voltage in a tetrode or pentode. Also, in the usual tetrode or pentode the plate current is substantially independent of plate voltage over the usual operating range- which means simply that the effective plate resistance of such tubes is relatively high. However, when the plate voltage falls below the normal operating range, the plate current falls sharply, while the screen current rises to such a value that the total cathode current remains substantially constant. Hence, the screen grid in a tetrode or pentode will almost invariably be damaged by excessive dissipation if the plate voltage is removed while the screen voltage is still being applied from a low -impedance source. Tubes 79 The current equations show how the total cathode current in triodes, tetrodes, and pentodes is a function of the potentials applied to the various electrodes. If only one electrode is positive with respect to the cathode (such as would be the case in a triode acting as a class A amplifier) all the cathode current goes to the plate. But when both screen and plate are positive in a tetrode or pentode, the cathode current divides between the two elements. Hence the screen current is taken from the total cathode current, while the balance goes to the plate. Further, if the control grid in a tetrode or pentode is operated at a positive potential the total cathode current is divided between all three elements which have a positive potential. In a tube which is receiving a large excitation voltage, it may be said that the control grid robs electrons from the output electrode during the period that the grid is positive, making it always necessary to limit the peak -positive excursion of the control grid. The Effect of Grid Current In general it may be stated that the amplification factor of tetrode and pentode tubes is a coefficient which is not of much use to the designer. In fact the amplification factor is seldom given on the design data sheets of such tubes. Its value is usually very high, due to the relatively high plate resistance of such tubes, but bears little relationship to the stage gain which actually will be obtained with such tubes. On the other hand, the grid-plate transconductance is the most important coefficient of pentode and tetrode tubes. Gain per stage can be computed directly when the Gm is known. The grid -plate transconductance of a tetrode or pentode tube can be calculated through use of the 'expression: Alp Coefficients of Tetrodes and Pentodes Gm = AE e with E5s and Es constant. The plate resistance of such tubes is of less importance than in the case of triodes, though it is often of value in determining the amount of damping a tube will exert upon the impedance in its plate circuit. Plate resistance is calculated from: AEp R v with Es and Esg constant. 4 -5 The Mixer and Converter Tubes superheterodyne receiver always in- www.americanradiohistory.com 80 Vacuum Tube Principles THE RADIO OSCILLATOR GRID - PLATE r_FSCREEN CAT NODE GRID METAL SPELL FILAMENT ` SUPPRESSOR AND SHELL Figure SIGNAL GRID Figure 20 LEAD INDUCTANCE GRID STRUCTURE OF 6SA7 The degenerative action of cathode lead inductance tends to reduce the effective grid-tocathode voltage with respect to the voltage available across the input tuned circuit. Cathode lead inductance also introduces undesirable coupling between the input and the out- CONVERTER TUBE eludes at least one stage for changing the frequency of the incoming signal to the fixed frequency of the main intermediate amplifier in the receiver. This frequency changing process is accomplished by selecting the beat -note difference frequency between a locally generated oscillation and the incoming signal frequency. If the oscillator signal is supplied by a separate tube, the frequency changing tube is called a mixer. Alternatively, the oscillation may be generated by additional elements within the frequency changer tube. In this case the frequency changer is commonly called a converter tube. Conversion Conductance The conversion conductance(Ge) is a coefficient of interest in the case of mixer or converter tubes, or of conventional triodes, tetrodes, or pentodes operating as frequency changers. The conversion conductance is the ratio of a change in the signal -grid voltage at the input frequency to a change in the output current at the converted frequency. Hence Gc in a mixer is essentially the same as transconductance in an amplifier, with the exception that the input signal and the output current are on different frequencies. The value of G, in conventional mixer tubes is from 300 to 1000 micromhos. The value of G, in an amplifier tube operated as a mixer is approximately 0.3 the Gm of the tube operated as an amplifier. The voltage gain of a mixer stage is equal to GCZL where ZL is the impedance of the plate load into which the mixer tube operates. The simplest mixer tube is the diode. The noise figure, or figure of merit, for a mixer of this type is not as good as that obtained with other more complex mixers; however, the diode is useful as a mixer in u -h -f and v -h -f equipment where low interelectrode capacities are vital to circuit operation. Since the diode impedance is The Diode Mixer 21 SHOWING THE EFFECT OF CATHODE put circuits. low, the local oscillator must furnish considerable power to the diode mixer. A good diode mixer has an overall gain of about 0.5. A triode mixer has better gain and a better noise figure than the diode mixer. At low frequencies, the gain and noise figure of a triode mixer closely approaches those figures obtained when the tube is used as an amplifier. In the u -h -f and v -h -f range, the efficiency of the triode mixer deteriorates rapidly. The optimum local oscillator voltage for a triode mixer is about 0.7 as large as the cutoff bias of the triode. Very little local oscillator power is required by a triode mixer. The Triode Mixer Pentode Mixers and Converter Tubes The most common multi grid converter tube for broadcast or shortwave use is the penta grid converter, typified by the 6SA7, 6SB7 -Y and 6BA7 tubes (figure 20). Operation of these converter tubes and pentode mixers will be covered in the Receiver Fundamentals Chapter. 4 -6 Electron Tubes at Very High Frequencies As the frequency of operation of the usual type of electron tube is increased above about 20 Mc., certain assumptions which are valid for operation at lower frequencies must be reexamined. First, we find that lead inductances from the socket connections to the actual elements within the envelope no longer are negligible. Second, we find that electron www.americanradiohistory.com HANDBOOK The transit time no longer may be ignored; an appreciable fraction of a cycle of input signal may be required for an electron to leave the cathode space charge, pass through the grid wires, and travel through the space between grid and plate. The effect of lead inductance is two -fold. First, as shown in figure 21, the combination of grid -lead inductance, gridcathode capacitance, and cathode lead inductance tends to reduce the effective grid- cathode signal voltage for a constant voltage at the tube terminals as the frequency is increased. Second, cathode lead inductance tends to introduce undesired coupling between the various elements within the tube. Tubes especially designed for v -h -f and u -h -f use have had their lead inductances minimized. The usual procedures for reducing lead inductance are: (1) using heavy lead conductors or several leads in parallel (examples are the 6SH7 and 6AK5), (2) scaling down the tube in all dimensions to reduce both lead inductances and interelectrode capacitances (examples are the 6AK5, 6F4, and other acorn and miniature tubes), and (3) the use of very low inductance extensions of the elements themselves as external connections (examples are lighthouse tubes such as the 2C40, oilcan tubes such as the 2C29, and many types of v -h -f transmitting tubes). Effects of Lead Inductance Effect of Transit Time When erated an electron tube is opat a frequency high enough that electron transit time between cathode and plate is an appreciable fraction of a cycle at the input frequency, several undesirable effects take place. First, the grid takes power from the input signal even though the grid is negative at all times. This comes about since the grid will have changed its potential during the time required for an electron to pass from cathode to plate. Due to interaction, and a resulting phase difference between the field associated with the grid and that associated with a moving electron, the grid presents a resistance to an input signal in addition to its normal "cold" capacitance. Further, as a result of this action, plate current no longer is in phase with grid voltage. An amplifier stage operating at a frequency high enough that transit time is appreciable: (a) Is difficult to excite as a result of grid loss from the equivalent input grid resistance, (b) Is capable of less output since transconductance is reduced and plate current is not in phase with grid voltage. The effects of transit time increase with the square of the operating frequency, and they Klystron 81 increase rapidly as frequency is increased above the value where they become just appreciable. These effects may be reduced by scaling down tube dimensions; a procedure which also reduces lead inductance. Further, transit-time effects may be reduced by the obvious procedure of increasing electrode potentials so that electron velocity will be increased. However, due to the law of electron motion in an electric field, transit time is increased only as the square root of the ratio of operating potential increase; therefore this expedient is of limited value due to other limitations upon operating voltages of small electron tubes. 4 -7 Special Microwave Electron Tubes Due primarily to the limitation imposed by transit time, conventional negative -grid electron tubes are capable of affording worthwhile amplification and power output only up to a definite upper frequency. This upper frequency limit varies from perhaps 100 Mc. for conventional tube types to about 4000 Mc. for specialized types such as the lighthouse tube. Above the limiting frequency, the conventional negative -grid tube no longer is practicable and recourse must be taken to totally different types of electron tubes in which electron transit time is not a limitation to operation. Three of the most important of such microwave tube types are the klystron, the magnetron, and the travelling wave tube. The klystron is a type of electron tube in which electron transit time is used to advantage, Such tubes comprise, as shown in figure 22, a cathode, a focussing electrode, a resonator connected to a pair of grids which afford velocity modulation of the electron beam (called the "buncher "), a drift space, and another resonator connected to a pair of grids (called the "catcher "). A collector for the expended electrons may be included at the end of the tube, or the catcher may also perform the function of electron collection. The tube operates in the following manner: The cathode emits a stream of electrons which is focussed into a beam by the focussing electrode. The stream passes through the buncher where it is acted upon by any field existing between the two grids of the buncher cavity. When the potential between the two grids is zero, the stream passes through without change in velocity. But when the potential between the two grids of the buncher is increasingly positive in the direction of electron The Power Klystron www.americanradiohistory.com 82 Vacuum TWO- CAVITY A conventional with a feedback two cavities so Tube Principles Figure 22 KLYSTRON OSCILLATOR two -cavity klystron is shown loop connected between the that the tube may be used as an oscillator. motion, the velocity of the electrons in the beam is increased. Conversely, when the field becomes increasingly negative in the direction of the beam (corresponding to the other half cycle of the exciting voltage from that which produced electron acceleration) the velocity of the electrons in the beam is decreased. When the velocity- modulated electron beam reaches the drift space, where there is no field, those electrons which have been sped up on one half -cycle overtake those immediately ahead which were slowed down on the other half- cycle. In this way, the beam electrons become bunched together. As the bunched groups pass through the two grids of the catcher cavity, they impart pulses of energy to these grids. The catcher grid -space is charged to different voltage levels by the passing electron bunches, and a corresponding oscillating field is set up in the catcher cavity. The catcher is designed to resonate at the frequency of the velocity- modulated beam, or at a harmonic of this frequency. In the klystron amplifier, energy delivered by the buncher to the catcher grids is greater than that applied to the buncher cavity by the input signal. In the klystron oscillator a feedback loop connects the two cavities. Coupling to either buncher or catcher is provided by small Loops which enter the cavities by way of concentric lines. The klystron is an electron -coupled device. When used as an oscillator, its output voltage is rich in harmonics. Klystron oscillators of various types afford power outputs ranging from less than I watt to many thousand watts. Operating efficiency varies between 5 and 30 per cent. Frequency may be shifted to some extent by varying the beam voltage. Tuning is THE RADIO Figure 23 REFLEX KLYSTRON OSCILLATOR A conventional reflex klystron oscillator of the type commonly used as o local oscillator in superheterodyne receivers operating above about 2000 Mc. is shown above. Frequency modulation of the output frequency of the oscillator, or o-f-c operation in a receiver, may be obtained by varying the negative voltage on the repeller electrode. carried on mechanically in some klystrons by altering (by means of knob settings) the shape of the resonant cavity. two -cavity klystron as described in the preceding paragraphs is primarily used as a transmitting device since quite reasonable amounts of power are made available in its output circuit. However, for applications where a much smaller amount of power is required-power levels in the milliwatt range for low -power transmitters, receiver local oscillators, etc., another type of klystron having only a single cavity is more frequently used. The theory of operation of the single- cavity klystron is essentially the same as the multi cavity type with the exception that the velocity- modulated electron beam, after having left the " buncher" cavity is reflected back into the area of the buncher again by a repeller electrode as illustrated in figure 23. The potentials on the various electrodes are adjusted to the value such that proper bunching of the electron beam will take place just as a particular portion of the velocity -modulated beam reenters the area of the resonant cavity. Since this type of klystron has only one circuit it can be used only as an oscillator and not as an amplifier. Effective modulation of the frequency of a single- cavity klystron for FM work can be obtained by modulating the repeller electrode voltage. The Reflex Klystron www.americanradiohistory.com The - HANDBOOK Magnetron The PLATE 83 MAGNET COIL 1 ANODE ANODE FIL GRID TERMINAL ANODE TERMINAL II` CATHODE / IL ANODE GLASS `PLATE 2 SEAL GLASS ENVELOPE O ANODE OR'D HEATER FILAMENT VOLTAGE EYELET SEAL LEAD TERMINAL EYELET TURULATiON lower Figure 24 CUTAWAY VIEW OF WESTERN ELECTRIC 416- B/6280 VHF PLANAR TRIODE TUBE The 416 -B, designed by the Bell Telephone Laboratories is intended for amplifier or frequency multiplier service in the 4000 me region. Employing grid wires having a diameter equal to fifteen wavelengths of light, 416 -B has a transconductance of 50,000. Spacing between grid and cathode is .0005', to reduce transit time effects. Entire tube is gold plated. The Magnetron magnetron is an oscillator tube normally The PLATE VOLTAGE Figure 25 SIMPLE MAGNETRON OSCILLATOR An external tank circuit is used with this type of magnetron oscillator for operation in the GLASS the FILAMENT s -h -f em- ployed where very high values of peak power or moderate amounts of average power are required in the range from perhaps 700 Mc. to 30,000 Mc. Special magnetrons were developed for wartime use in radar equipments which had peak power capabilities of several million watts (megawatts) output at frequencies in the vicinity of 3000 Mc. The normal duty cycle of operation of these radar equipments was approximately 1 /10 of one per cent (the tube operated about 1 /1000 of the time and rested for the balance of the operating period) so that the average power output of these magnetrons was in the vicinity of 1000 watts. u -h -f ronge. In its simplest form the magnetron tube is a filament -type diode with two half-cylindrical plates or anodes situated coaxially with respect to the filament. The construction is illustrated in figure 25A. The anodes of the magnetron are connected to a resonant circuit as illustrated on figure 25B. The tube is surrounded by an electromagnet coil which, in turn, is connected to a low -voltage d -c energizing source through a rheostat R for controlling the strength of the magnetic field. The field coil is oriented so that the lines of magnetic force it sets up are parallel to the axis of the electrodes. Under the influence of the strong magnetic field, electrons leaving the filament are deflected from their normal paths and move in circular orbits within the anode cylinder. This effect results in a negative resistance which sustains oscillations. The oscillation frequency is very nearly the value determined by L and C. In other magnetron circuits, the frequency may be governed by the electron rotation, no external tuned circuits being employed. Wavelengths of less than 1 centimeter have been produced with such circuits. More complex magnetron tubes employ no external tuned circuit, but utilize instead one or more resonant cavities which are integral with the anode structure. Figure 26 shows a magnetron of this type having a multi -cellular www.americanradiohistory.com 84 Vacuum Tube Principles - RADIO THE WAVE GUIDE CATNODE LEAOE WAVE GU IDE OUTPUT INPUT ELECTRON BEAM MAGNETRON PE MANE NT CTNODE MAGNET ANODE ESSASS IIII rT TING Outryt NODE BLOC iii1!,;,'+ Figure 26 MODERN MULTI- CAVITY MAGNETRON Illustrated is an external -anode strapped magnetron of the type commonly used in radar equipment for the 10 -cm. range. A permanent magnet of the general type used with such a magnetron Is shown in the right -hand portion of the drawing, with the magnetron in place between the pole pieces of the magnet. anode of eight cavities. It will be noted, also, that alternate cavities (which would operate at the same polarity when the tube is oscillating) are strapped together. Strapping was found to improve the efficiency and stability of high power radar magnetrons. In most radar applications of magnetron oscillators a powerful permanent magnet of controlled characteristics is employed to supply the magnetic field rather than the use of an electromagnet. The Travelling Wave Tube Travelling Wave Tube (figure 27) consists of a helix located within an evacuated envelope. Input and output terminations are affixed to each end of the helix. An electron beam passes through the helix and interacts with a wave travelling along the helix to produce broad band amplification at microwave frequencies. When the input signal is applied to the gun end of the helix, it travels along the helix wire at approximately the speed of light. However, the signal velocity measured along the axis of the helix is considerably lower. The electrons emitted by the cathode gun pass axially through the helix to the collector, located at the output end of the helix. The average velocity of the electrons depends upon the potential of the collector with respect to the cathode. When the average velocity of the electrons is greater than the velocity of the helix wave, the electrons become crowded together in the various regions of retarded field, where they impart energy to the helix wave. A power gain of 100 or more may be produced by this tube. 4 -8 The The Cathode -Ray Tube The Cathode -Ray Tube cathode -ray The is a tube special type of ANODE COLLECTOR Figure 27 THE TRAVELLING WAVE TUBE Operation of this tube is the result of inter. action between the electron beam and wave travelling along the helix. electron tube which permits the visual observation of electrical signals. It may be incorporated into an oscilloscope for use as a test instrument or it may be the display device for radar equipment or a television receiver. cathode -ray tube always includes an electron gun for producing a stream of electrons, a grid for controlling the intensity of the electron beam, and a luminescent screen for converting the impinging electron beam into visible light. Such a tube always operates in conjunction with either a built -in or an external means for focussing the electron stream into a narrow beam, and a means for deflecting the electron beam in accordance with an electrical signal. The main electrical difference between types of cathode -ray tubes lies in the means employed for focussing and deflecting the electron beam. The beam may be focussed and/or deflected either electrostatically or magnetically, since a stream of electrons can be acted upon either by an electrostatic or a magnetic field. In an electrostatic field the electron beam tends to be deflected toward the positive termination of the field (figure 28). In a magnetic field the stream tends to be deflected at right angles to the field. Further, an electron beam tends to be deflected so that it is normal (perpendicular) to the equipotential lines of an electrostatic field- and it tends to be deflected so that it is parallel to the lines of force in a magnetic field. Large cathode -ray tubes used as kinescopes in television receivers usually are both focused and deflected magnetically. On the other hand, the medium -size CR tubes used in oscilloscopes and small television receivers usually are both focused and deflected electrostatically. But CR tubes for special applications may be focused magnetically and deflected electrostatically or vice versa. There are advantages and disadvantages to Operation of the CRT A www.americanradiohistory.com HANDBOOK The - NOR ACCELERATING ANODE IN) SASE NEATE C Lr. LEG T DE IFI I LONTAL DEFLECTION ICI OA r1 -ADUADAG SECONDARY COATING -1 è. CLCCTIINNEEAu RUORESCENT CONTROL ACCELERATI ANODE (A) GRID IG CATNODE (10 SCREEN ZATNODE -VERTICAL DEFLECTION PLATES IS) Figure 28 TYPICAL ELECTROSTATIC CATHODE -RAY TUBE both types of focussing and deflection. However, it may be stated that electrostatic deflection is much better than magnetic deflection when high -frequency waves are to be displayed on the screen; hence the almost universal use of this type of deflection for oscillographic work. But when a tube is operated at a high value of accelerating potential so as to obtain a bright display on the face of the tube as for television or radar work, the use of magnetic deflection becomes desirable since it is relatively easier to deflect a high -velocity electron than electrostatically. beam magnetically However, an ion trap is required with magnetic deflection since the heavy negative ions emitted by the cathode are not materially deflected by the magnetic field and hence would burn an "ion spot" in the center of the luminescent screen. With electrostatic deflection the heavy ions are deflected equally as well as the electrons in the beam so that an ion spot is not formed. Construction of The construction of a typical Electrostatic CRT electrostatic- focus, electrostatic- deflection cathode-ray tube is illustrated in the pictorial diagram of figure 28. The indirectly heated cathode K releases free electrons when heated by the enclosed filament. The cathode is surrounded by a cylinder G, which has a small hole in its front for the passage of the electron stream. Although this element is not a wire mesh as is the usual grid, it is known by the same name because its action is similar: it controls the electron stream when its negative potential is varied. Next in order, is found the first accelerating anode, H, which resembles another disk or cylinder with a small hole in its center. This electrode is run at a high or moderately high positive voltage, to accelerate the electrons towards the far end of the tube. The focussing electrode, F, is a sleeve which usually contains two small disks, each with a small hole. After leaving the focussing electrode, the electrons pass through another accelerating Cathode Ray Tube 85 anode, A, which is operated at a high positive potential. In some tubes this electrode is operatcd at a higher potential than the first accelerating electrode, H, while in other tubes both accelerating electrodes are operated at the same potential. The electrodes which have been described this point constitute the electron gun, which produces the free electrons and focusses them into a slender, concentrated, rapidly traveling stream for projecting onto the viewing screen. up to Electrostatic Deflection To make the tube useful, means must be provided for deflecting the electron beam along two axes at right angles to each other. The more corn mon tubes employ electrostatic deflection plates, one pair to exert a force on the beam in the vertical plane and one pair to exert a force in the horizontal plane. These plates are designated as B and C in figure 28. Standard oscilloscope practice with small cathode -ray tubes calls for connecting one of the B plates and one of the C plates together and to the high voltage accelerating anode. With the newer three -inch tubes and with five inch tubes and larger, all four deflecting plates are commonly used for deflection. The positive high voltage is grounded, instead of the negative as is common practice in amplifiers, etc., in order to permit operation of the deflecting plates at a d-c potential at or near ground. An Aquadag coating is applied to the inside of the envelope to attract any secondary electrons emitted by the flourescent screen. In the average electrostatic -deflection CR tube the spot will be fairly well centered if all four deflection plates are returned to the po- tential of the second anode (ground). How ever, for accurate centering and to permit moving the entire trace either horizontally or vertically to permit display of a particular waveform, horizontal and vertical centering controls usually are provided on the front of the oscilloscope. After the spot is once centered, it is necessary only to apply a positive or negative voltage (with respect to ground) to one of the ungrounded or "free" deflector plates in order to move the spot. If the voltage is positive with respect to ground, the beam will be attracted toward that deflector plate, while if negative the beam and spot will be repulsed. The amount of deflection is directly proportional to the voltage (with respect to ground) that is applied to the free electrode. With the larger- screen higher -voltage tubes it becomes necessary to place deflecting voltage on both horizontal and both vertical plates. This is done for two reasons: First, the amount of deflection voltage required by the high- www.americanradiohistory.com Vacuum 86 Tube FIRST ANODE \ THE DEFLECTION COILS . 11111 SECOND (ADYADA.,Ï _ /CONTROL GRID RADIO T-TERMINAL FOCUS COIL BASE Principles ® 10 _ _ 1tI1.-o R_iirr- -v ECEETFOÑR[AM FLUORESCENT SCREEN - ID) /CATHODE (R) A R Figure 29 TYPICAL ELECTROMAGNETIC CATHODE -RAY TUBE voltage tubes is so great that a transmitting tube operating from a high voltage supply would be required to attain this voltage without distortion. By using push -pull deflection with two tubes feeding the deflection plates, the necessary plate supply voltage for the deflection amplifier is halved. Second, a certain amount of de- focussing of the electron stream is always present on the extreme excursions in deflection voltage when this voltage is applied only to one deflecting plate. When the deflecting voltage is fed in push -pull to both deflecting plates in each plane, there is no defocussing because the average voltage acting on the electron stream is zero, even though the net voltage (which causes the deflection) acting on the stream is twice that on either plate. The fact that the beam is deflected by a magnetic field is important even in an oscilloscope which employs a tube using electrostatic deflection, because it means that precautions must be taken to protect the tube from the transformer fields and sometimes even the earth's magnetic field. This normally is done by incorporating a magnetic shield around the tube and by placing any transformers as far from the tube as possible, oriented to the position which produces minimum effect upon the electron stream. The electromagnetic cathode -ray tube allows greater definition than does the electrostatic tube. Also, electromagnetic definition has a number of advantages when a rotating radial sweep is required to give polar indications. The production of the electron beam in an electromagnetic tube is essentially the same as in the electrostatic tube. The grid structure is similar, and controls the electron beam in an identical manner. The elements of a typical electromagnetic tube are shown in figure 29. The focus coil is wound on an iron core which may be moved along the neck of the tube to focus the electron beam. For final adjustment, Construction of Electro- magnetic CRT .1m Jew .1n1R Figure 30 Two pairs of coils arranged for electromagnetic deflection in two directions. the current flowing in the coil may be varied. A second pair of coils, the deflection coils are mounted at right angles to each other around the neck of the tube. In some cases, these coils can rotate around the axis of the tube. Two anodes are used for accelerating the electrons from the cathode to the screen. The second anode is a graphite coating (Aquadag) on the inside of the glass envelope. The function of this coating is to attract any secondary electrons emitted by the flourescent screen, and also to shield the electron beam. In some types of electromagnetic tubes, a first, or accelerating anode is also used in addition to the Aquadag. Electromagnetic Deflection magnetic field will deflect an electron beam in a direction which is at right angles to both the direction of the field and the direction of motion of the beam. In the general case, two pairs of deflection coils are used (figure 30). One pair is for horizontal deflection, and the other pair is for vertical deflection. The two coils in a pair are connected in series and are wound in such directions that the magnetic field flows from one coil, through the electron beam to the other coil. The force exerted on the beam by the field moves it to any point on the screen by application of the proper currents to these A coils. The human eye retains an image for about one - sixteenth second after viewing. In a CRT, the spot can be moved so quickly that a series of adjacent spots can be made to appear as a line, if the beam is swept over the path fast enough. As The Trace www.americanradiohistory.com HANDBOOK Gas long as the electron beam strikes in a given place at least sixteen times a second, the spot will appear to the human eye as a source of continuous light with very little flicker. - least five types of luminescent screen materials are commonly available on the various types of CR tubes commercially available. These screen materials are called phosphors; each of the five phosphors is best suited to a particular type of application. The P -1 phosphor, which has a green flourescence with medium persistence, is almost invariably used for oscilloscope tubes for visual observaScreen Materials "Phosphors" At tion. The P -4 phosphor, with white fluorescence and medium persistence, is used on television viewing tubes ( "Kinescopes "). The P -5 and P -11 phosphors, with blue fluorescence and very short persistence, are used primarily in oscilloscopes where photographic recording of the trace is to be obtained. The P -7 phosphor, which has a blue flash and a long -persistence greenish -yellow persistence, is used primarily for radar displays where retention of the image for several seconds after the initial signal display is required. 4 -9 Gas Tubes The space charge of electrons in the vicinity of the cathode in a diode causes the plate -tocathode voltage drop to be a function of the current being carried between the cathode and the plate. This voltage drop can be rather high when large currents are being passed, causing a considerable amount of energy loss which shows up as plate dissipation. Mercury Vapor Tubes Tubes 87 Mercury -vapor tubes, although very widely used, have the disadvantage that they must be operated within a specific temperature range (25° to 70°C.) in order that the mercury vapor pressure within the tube shall be within the proper range. If the temperature is too low, the drop across the tube becomes too high causing immediate overheating and possible damage to the elements. If the temperature is too high, the vapor pressure is too high, and the voltage at which the tube will "flash back" is lowered to the point where destruction of the tube may take place. Since the ambient temperature range specified above is within the normal room temperature range, no trouble will be encountered under normal operating conditions. However, by the substitution of xenon gas for mercury it is possible to produce a rectifier with characteristics comparable to those of the mercury -vapor tube except that the tube is capable of operating over the range from approximately -70° to 90° C. The 3B25 rectifier is an example of this type of tube. If a grid is inserted between the cathode and plate of a mercury -vapor gaseous- conduction rectifier, a negative potential placed upon the added element will increase the plate -to- cathode voltage drop required before the tube will ionize or "fire." The potential upon the control grid will have no effect on the plate -to- cathode drop after the tube has ionized. However, the grid voltage may be adjusted to such a value that conduction will take place only over the desired portion of the cycle of the a-c voltage being impressed upon the plate of the rectifier. Thyratron Tubes Voltage Regulator In a glow -discharge gas tube Tubes the voltage drop across the electrodes remains constant The negative space charge can be neutralized by the presence of the proper density of positive ions in the space between the cathode and anode. The positive ions may be obtained by the introduction of the proper amount of gas or a small amount of mercury into the envelope of the tube. Then the voltage drop across the tube reaches the ionization potential of the gas or mercury vapor, the gas molecules will become ionized to form positive ions. The positive ions then tend to neutralize the space charge in the vicinity of the cathode. The voltage drop across the tube then remains constant at the ionization potential of the gas up to a current drain equal to the maximum emission capability of the cathode. The voltage drop varies between 10 and 20 volts, depending upon the particular gas employed, up to the maximum current rating of the tube. Action of Positive Ions over a wide range of current passing through the tube. This property exists because the degree of ionization of the gas in the tube varies with the amount of current passing through the tube. When a large current is passed, the gas is highly ionized and the internal impedance of the tube is low. When a small current is passed, the gas is lightly ionized and the internal impedance of the tube is high. Over the operating range of the tube, the product (IR) of the current through the tube and the internal impedance of the tube is very nearly constant. Examples of this type of tube are VR -150, VR -105 and the old 874. Vacuum tubes are grouped into three major classifications: commercial, ruggedized, and premium (or reliable). Any one of these three groups may also be further classified for Vacuum Tube Classification www.americanradiohistory.com 88 Tube Vacuum THE Principles 100- military duty (JAN classification). To qualify for JAN classification, sample lots of the particular tube must have passed special qualification tests at the factory. It should not be construed that a JAN-type tube is better than a commercial tube, since some commercial tests and specifications are more rigid than the corresponding JAN specifications. The JAN -stamped tube has merely been accepted under a certain set of conditions for military service. IP=2.5IAA. eo 60 L 0 20 0 0 10 20 EP Ruggedized or Premium Tubes Radio tubes are being used in increasing numbers for industrial applications, such as computing and control machinery, and in aviation and marine equipment. When a tube fails in a home radio receiver, it is merely inconvenient, but a tube failure in industrial applications may bring about stoppage of some vital process, resulting in financial loss, or even danger to life. To meet the demands of these industrial applications, a series of tubes was evolved incorporating many special features designed to ensure a long and pre- determined operating life, and uniform characteristics among similar tubes. Such tubes are known as ruggedized or premium tubes. Early attempts to select reTRIODE PLATE , `FLUORESCENT ANODE TRIODE GRID RAY CONTROL ELECTRODE CATHODES Figure 31 SCHEMATIC REPRESENTATION OF "MAGIC EYE" TUBE liable specimens of tubes from ordinary stock tubes proved that in the long run the selected tubes were no better than tubes picked at random. Long life and ruggedness had to be built into the tubes by means of proper choice and 100% inspection of all materials used in the tube, by critical processing inspection and assembling, and by conservative ratings of the tube. Pure tungsten wire is used for heaters in preference to alloys of lower tensile strength. Nickel tubing is employed around the heater wires at the junction to the stem wires to reduce breakage at this point. Element structures are given extra supports and bracing. Finally, all tubes are given a 50 hour test run under full operating conditions to eliminate early failures. When operated within their ratings, ruggedized or premium tubes should provide a life well in excess of 10,000 hours. Ruggedized tubes will withstand severe impact shocks for short periods, and will RADIO 30 40 VOLTS) 50 60 Figure 32 AMPLIFICATION FACTOR OF TYPICAL MODE TUBE DROPS RAPIDLY AS PLATE VOLTAGE IS DECREASED BELOW 20 VOLTS operate under conditions of vibration for many hours. The tubes may be identified in many cases by the fact that their nomenclature includes a "W" in the type number, as in 807W, 5U4W, etc. Some ruggedized tubes are included in the "5000" series nomenclature. The 5654 is a ruggedized version of the 6AK5, the 5692 is a ruggedized version of the 6SN7, etc. 4 -10 lectron Miscellaneous Tube Types electron -ray tube or magic eye contains two sets of elements, one of which is a triode amplifier and the other a cathode -ray indicator. The plate of the triode section is internally connected to the ray- control electrode (figure 31), so that as the plate voltage varies in accordance with the applied signal the voltage on the ray -control electrode also varies. The ray -control electrode is a metal cylinder so placed relative to the cathode that it deflects some of the electrons emitted from the cathode. The electrons which strike the anode cause it to fluoresce, or give off light, so that the deflection caused by the ray -control electrode, which prevents electrons from striking part of the anode, produces a wedge- shaped electrical shadow on the fluorescent anode. The size of this shadow is determined by the voltage on the ray -electrode. When this electrode is at the same potential as the fluorescent anode, the shadow disappears; if the ray -electrode is less positive than the anode, a shadow appears the width of which is proportional to the voltage on the ray -electrode. Magic eye tubes may be used as tuning indicators, and as balance indicators in electrical bridge circuits. If the angle of shadow is calibrated, the eye tube may be used as a voltmeter where rough measurements suffice. E The Ray Tubes www.americanradiohistory.com Miscellaneous HANDBOOK 89 Ecz (VOLTS) Ec1=,z.ev Figure 33 CHARACTERISTIC CURVES OF 12AK5 SFACE- CHARGE TRIODE Controlled Warm -up Tubes Series heater strings are employed in ac -dc radio receivers and television sets to reduce the cost, size, and weight of the equipment. Voltage surges of great magnitude occur in series operated filaments because of variations in the rate of warm -up of the various tubes. As the tubes warm up, the heater resistance changes. This change is not the same between tubes of various types, or even between tubes same type made by different manufacturers. Some 6 -volt tubes show an initial surge as high as 9 -volts during warm -up, while slow -heating tubes such as the 25BQ6 are underheated during the voltage surge on the 6 -volt tubes. the of Standardization of heater characteristics in a new group of cubes designed for series heater strings has eliminated this trouble. The new tubes have either 600 ma. or 400 ma. heaters, with a controlled warm -up time of approximately 11 seconds. The 5U8, 6CG7, and 12BH7 -A are examples of controlled warm -up tubes. Introduction of the 12 -volt ignition system in American automobiles Potential has brought about the design of a Tubes series of tubes capable of operation with a plate potential of 12 -14 volts. Standard tubes perform poorly at low plate potentials, as the amplification factor of the tube drops rapidly as the plate voltage is decreased (figure 32). Contact potential effects, and change of characteristics with variations of filament voltage combine to make operation at low plate potentials even more Low Plate erratic. By employing and by altering the electrode geometry a series of low voltage tubes has been developed by Tung -Sol that effectively perform with all electrodes energized by a 12 -volt system. With a suitable power output transistor, this makes possible an automobile radio without a vibrator power supply. A special space- charge tube (12K5) has been developed that delivers 40 milliwatts of audio power with a 12 volt plate supply (figure 33). The increased number of imported radios and high- fidelity equipment have brought many foreign vacuum tubes into the United States. Many of these tubes are comparable to, or interchangeable with standard American tubes. A complete listing of the electrical characteristics and hase connection diagrams of all general purpose tubes made in all tube -producing countries outside the "Iron Curtain" is contained in the Radio Tube Vade Mecum (World's Radio Tubes) available at most larger radio parts jobbers or by mail from the publishers of this Handbook. The Equivalent Tubes Vade Mecum (World's Equivalent Tubes) gives all replacement tubes for a given type, both exact and near -equivalents (with points of difference detailed). (Data on TV and special purpose tubes if needed is contained in a companion volume Television Tubes Vade Mecum). Foreign Tubes special processing techniques www.americanradiohistory.com CHAPTER FIVE Transistors and Semi -Conductors One of the earliest detection devices used in radio was the galena crystal, a crude example of a semiconductor. More modern examples of semiconductors are the copper - 5 -1 It has been previously stated that the electrons in an element having a large atomic number are grouped into rings, each ring having a definite number of electrons. Atoms in which these rings are completely filled are called inert gases, of which helium and argon are examples. All other elements have one or more incomplete rings of electrons. If the incomplete ring is loosely bound, the electrons may be easily removed, the element is called metallic, and is a conductor of electric current. If the incomplete ring is tightly bound, with only a few missing electrons, the element is called non - metallic and is an insulator of electric current. Germanium and silicon fall between these two sharply defined groups, and exhibit both metallic and non -metallic characteristics. Pure germanium or silicon may be considered to be a good insulator. The addition of certain impurities in carefully controlled amounts to the pure germanium will alter the conductivity of the material. In addition, the choice of the impurity can change the direction of conductivity through the crystal, some impurities increasing conductivity to positive voltages, and others increasing conductivity to negative voltages. oxide rectifier, the selenium rectifier and the germanium diode. All of these devices offer the interesting property of greater resistance to the flow of electrical current in one direction than in the opposite direction. Typical conduction curves for these semiconductors are shown in Figure 1. The copper oxide rectifier action results from the function of a thin film of cuprous oxide formed upon a pure copper disc. This film offers low resistance for positive voltages, and high resistance for negative voltages. The same action is observed in selenium rectifiers, where a film of selenium is deposited on an iron surface. s 1 1N3 CD0IIT*L DIODI O 1 TYPICAL STATIC CHARACTERISTICS 00 w w o I. I 1.1 I.1 -00 -.0 -SO - -20 0 0 5 -2 2 Mechanism of Conduction As indicated by their name, semiconductors are substances which have a conductivity intermediate between the high values observed for metals and the low values observed for insulating materials. The mechanism of conduction in semiconductors is different from that a VOLTS Figure Atomic Structure of Germanium and Silicon lA TYPICAL CHARACTERISTIC CURVE OF SEMI -CONDUCTOR DIODE 90 www.americanradiohistory.com Transistors observed in metallic conductors. There exist in semiconductors both negatively charged electrons and positively charged particles, called holes, which behave as though they had a positive electrical charge equal in magnitude to the negative electrical charge on the electron. These holes and electrons drift in an electrical field with a velocity which is proportional to the field itself: SCHEMATIC REPRESENTATION VAN where VAN E _ -1=011 ANODES Color CATB000 BEOI M11 ¡ l. Calorr Bao ft -- Wrba TUBE. GERMANIUM. SILICON AND SELENIUM DIODES Figure COMMON DIODE AND MARKINGS IN ABOVE 1 91 -B COLOR CODES ARE SHOWN CHART = µnE = drift velocity of hole = magnitude of electric field = mobility of hole In an electric field the holes will drift in a direction opposite to that of the electron and with about one-half the velocity, since the hole mobility is about one -half the electron mobility. A sample of a semiconductor, such as germanium or silicon, which is both chemically pure and mechanically perfect will contain in it approximately equal numbers of holes and electrons and is called an intrinsic semiconductor. The intrinsic resistivity of the semiconductor depends strongly upon the temperature, being about 50 ohm /cm. for germanium at room temperature. The intrinsic resistivity of silicon is about 65,000 ohm /cm. at the same temperature. If, in the growing of the semiconductor crystal, a small amount of an impurity, such as phosphorous, arsenic or antimony is included in the crystal, each atom of the impurity contributes one free electron. This electron is available for conduction. The crystal is said to be doped and has become electron- conductPe- Nb JUNCTION PLASTIC CASE b-PC JUNCTION P - TYPE N- TYPE GERMANIUM CRYSTAL LAYER GERMANIUM CRYSTAL LAYER COLLECTOR EMITTER LI Nb P .MS' BASE CONNECTION SMALL 3 -PIN BASE L Jw y1! o o Pt 4- Z .320 ASE CONNECTION EMITTER COLLECTOR Figure 2A CUT -AWAY VIEW OF JUNCTION TRANSISTOR, SHOWING PHYSICAL ARRANGEMENT Figure 2B PICTORIAL EQUIVALENT OF P -N -P JUNCTION TRANSISTOR www.americanradiohistory.com SIGN Z 92 THE RADIO Transistors and Semi- Conductors EMITTER COLLECTOR EMITTER COLLECTOR BASE-I BISE TRANSISTOR OR POINT CONTACT TRANSISTOR N-P-N TRANSISTOR P -N -P Figure 4 ELECTRICAL SYMBOLS FOR TRANSISTORS Figure 3 CONSTRUCTION DETAIL OF A POINT CONTACT TRANSISTOR ing in nature and is called N (negative) type germanium. The impurities which contribute electrons are called donors. N -type germanium has better conductivity than pure germanium in one direction, and a continuous stream of electrons will flow through the crystal in this direction as long as an external potential of the correct polarity is applied across the crystal. Other impurities, such as aluminum, gallium or indium add one hole to the semiconducting crystal by accepting one electron for each atom of impurity, thus creating additional holes in the semiconducting crystal. The material is now said to be hole- conducting, or P (positive) type germanium. The impurities which create holes are called acceptors. P -type germanium has better conductivity than pure germanium in one direction. This direction is opposite to that of the N -type material. Either the N -type or the P -type germanium is called extrinsic conducting type. The doped materials have lower resistivities than the pure materials, and doped semiconductors in the resistivity range of .01 to 10 ohm /cm. are normally used in the production of transistors. 5-3 The Transistor In the past few years an entire new technology has been developed for the application of certain semiconducting materials in production of devices having gain properties. These gain properties were previously found only in vacuum tubes. The elements germanium and silicon are the principal materials which exhibit the proper semiconducting properties permitting their application in the new amplifying devices called transistors. However, other semiconducting materials, including the compounds indium antimonide and lead sulfide have been used experimentally in the production of transistors. Types of Transistors There are two basic types of transistors, the point-contact type and the junction type (figure 2) . Typical construction detail of a pointcontact transistor is shown in Figure 3, and the electrical symbol is shown in Figure 4. The emitter and collector electrodes make contact with a small block of germanium, called the base. The base may be either N -type or P -type germanium, and is approximately .05" long and .03" thick. The emitter and collector electrodes are fine wires, and are spaced about .005" apart on the germanium base. The complete assembly is usually encapsulated in a small, plastic case to provide ruggedness and to avoid contaminating effects of the atmosphere. The polarity of emitter and collector voltages depends upon the type of germanium employed in the base, as illustrated in figure 4. The junction transistor consists of a piece of either N -type or P -type germanium between two wafers of germanium of the opposite type. Either N -P -N or P -N -P transistors may be made. In one construction called the grown crystal process, the original crystal, grown from molten germanium or silicon, is created in such a way as to have the two closely spaced junctions imbedded in it. In the other construction called the fusion process, the crystals are grown so as to make them a single conductivity type. The junctions are then produced by fusing small pellets of special metal alloys into minute plates cut from the original crystal. Typical construction detail of a junction transistor is shown in figure 2A. The electrical schematic for the P -N -P junction transistor is the same as for the pointcontact type, as is shown in figure 4. Transistor Action Presently available types of transistors have three essential actions which collectively are called transistor action. These are: minority carrier injection, transport, and collection. Figure 2B shows a simplified drawing of a P-N -P junction -type transistor, which can illustrate this www.americanradiohistory.com HANDBOOK Transistors collective action. The P -N -P transistor consists of a piece of N -type germanium on opposite sides of which a layer of P -type material has been grown by the fusion process. Terminals are connected to the two P- sections and to the N -type base. The transistor may be considered as two P -N junction rectifiers p!aced in close juxaposition with a semi conduction crystal coupling the two rectifiers together. The left -hand terminal is biased in the forward (or conducting) direction and is called the emitter. The right -hand terminal is biased in the back (or reverse) direction and is called the collector The operating potentials are chosen with respect to the base terminal, which may or may not be grounded. If an N -P -N transistor is used in place of the P -N -P, the operating potentials are reversed. The P. Nb junction on the left is biased in the forward direction and holes from the P region are injected into the Nb region, producing therein a concentration of holes substantially greater than normally present in the material. These holes travel across the base region towards the collector, attracting neighboring electrons, finally increasing the available supply of conducting electrons in the collector loop. As a result, the collector loop possesses lower resistance whenever the emitter circuit is in operation. In junction transistors this charge transport is by means of diffusion wherein the charges move from a region of high concentration to a region of lower concentration at the collector. The collector, biased in the opposite direction, acts as a sink for these holes, and is said to collect them. It is known that any rectifier biased in the forward direction has a very low internal impedance, whereas one biased in the back direction has a very high internal impedance. Thus, current flows into the transistor in a low impedance circuit, and appears at the output as current flowing in a high impedance circuit. The ratio of a change in collector current to a change in emitter current is called the current amplification, or alpha: - a = ie = current amplification = change in collector current i. = change in emitter current where a ie Values of alpha up to 3 or so may be obtained in commercially available point- contact transistors, and values of alpha up to about 0.95 are obtainable in junction transistors. 93 Alpha Cutoff The alpha cutoff frequency of a transistor is that frequency at which the grounded base current gain has decreased to 0.7 of the gain obtained at 1 kc. For audio transistors, the alpha cutoff frequency is in the region of 0.7 Mc. to 1.5 Mc. For r -f and switching transistors, the alpha cutoff frequency may be 5 Mc. or higher. The upper frequency limit of operation of the transistor is determined by the small but finite time it takes the majority carrier to move from one electrode to another. Frequency Drift Transistors As previously noted, the signal current in a conventional transistor is transmitted across the base region by a diffusion process. The transit time of the carriers across this region is, therefore relatively long. RCA has developed a technique for the manufacture of transistors which does not depend upon diffusion for transmission of the signal across the base region. Transistors featuring this new process are known as drift transistors. Diffusion of charge carriers across the base region is eliminated and the carriers are propelled across the region by a "built in electric field. The resulting reduction of transit time of the carrier permits drift transistors to be used at much higher frequencies than transistors of conventional design. The "built in" electric field is in the base region of the drift transistor. This field is achieved by utilizing an impurity density which varies from one side of the base to the other. The impurity density is high next to the emitter and low next to the collector. Thus, there are more mobile electrons in the region near the emitter than in the region near the collector, and they will try to diffuse evenly throughout the base. However, any displacement of the negative charge leaves a positive charge in the region from which the electrons came, because every atom of the base material was originally electrically neutral. The displacement of the charge creates an electric field that tends to prevent further electron diffusion so that a condition of equilibrium is reached. The direction of this field is such as to prevent electron diffusion from the high density area near the emitter to the low density area near the collector. Therefore, holes entering the base will be accelerated from the emitter to the collector by the electric field. Thus the diffusion of charge carriers across the base region is augmented by the built -in electric field. A potential energy diagram for a drift transistor is shown in figure 5. www.americanradiohistory.com 94 \ % ri_ \ :i1, r_=/ THE RADIO Transistors and Semi -Conductors DECREASING POTENTIAL ENERGY OF S MAJORITY CARRIER h WW 7 CC 6 34 REGION u I EMITTER CASE REGION I I ION I COLLECTOR REGION I Ifi/Ciioill111E.1 IMTINEE:011101 o1 l'/. '/_!= S DRIFT u M IMI . ,1\I/I IIIPAlf I/ICJ 1 10 I I DISTANCE Figure 20 s0 40 50 COLLECTOR VOLTS 0 5 POTENTIAL ENERGY DIAGRAM DRIFT TRANSISTOR (2N247) FOR 5 -4 _Aiii/1111 Transistor Characteristics The transistor produces results that may be comparable to a vacuum tube, but there is a basic difference between the two devices. The vacuum tube is a voltage controlled device whereas the transistor is a current controlled device. A vacuum tube normally operates with its grid biased in the negative or high resistance direction, and its plate biased in the positive or low resistance direction. The tube conducts only by means of electrons, and has its conducting counterpart in the form of the N -P -N transistor, whose majority carriers are also electrons. There is no vacuum tube equivalent of the P -N -P transistor, whose majority carriers are holes. The biasing conditions stated above provide the high input impedance and low output impedance of the vacuum tube. The transistor is biased in the positive or low resistance direction in the emitter circuit, and in the negative, or high resistance direction in the collector circuit resulting in a low input impedance and a high output impedance, contrary to and opposite from the vacuum tube. A comparison of point-contact transistor characteristics and vacuum tube characteristics is made in figure 6. The resistance gain of a transistor is expressed as the ratio of output resistance to input resistance. The input resistance of a typical transistor is low, in the neighborhood of 300 ohms, while the output resistance is relatively high, usually over 20,000 ohms. For a point- contact transistor, the resistance gain is usually over 60. The voltage gain of a transistor is the product of alpha times the resistance gain, and for a point- contact transistor is of the FAlBM!lPA 25 so 75 100 125 ISO 175 200 PLATE VOLTS Figure 6 COMPARISON OF POINT -CONTACT TRANSISTOR AND VACUUM TUBE CHARACTERISTICS order of 3 X 60 = 180. A junction transistor which has a value of alpha less than unity nevertheless has a resistance gain of the order of 2000 because of its extremely high output resistance, and the resulting voltage gain is about 1800 or so. For both types of transistors the power gain is the product of alpha squared times the resistance gain and is of the order of 400 to 500. The output characteristics of the junction transistor are of great interest. A typical example is shown in figure 7. It is seen that the junction transistor has the characteristics of an ideal pentode vacuum tube. The collector current is practically independent of the collector voltage. The range of linear operation extends from a minimum voltage of about 0.2 volts up to the maximum rated collector voltage. A typical load line is shown, which illustrates the very high load impedance that would be required for maximum power transfer. A grounded emitter circuit is usually used, since the output impedance is not as high as when a grounded base circuit is used. www.americanradiohistory.com HANDBOOK Transistor Characteristics d 95 le COLLECTOR EMITTER CASE VALUES OF THE EQUIVALENT CIRCUIT POINT- CONTACT -1 0 -0.S 0 +5 +10 + 5 +20 ISTOR +25 COLLECTOR VOLTS Vs..iMA VC15V.) The output characteristics of a typical point contact transistor are shown in figure 6. The pentode characteristics are less evident, rind the output impedance is much lower, with the range of linear operation extending down to a collector voltage of 2 or 3. Of greater practical interest, however, is the input characteristic curve with short -circuited, or nearly shortcircuited input, as shown in figure 8. It is this point -contact transistor characteristic of having a region of negative impedance that lends the unit to use in switching circuits. The transistor circuit may be made to have two, one or zero stable operating points, depending upon the bias voltages and the load impedance used. Equivalent Circuit of a Transistor As is known from net - work theory, the small signal performance of any device in any network can be represented by means of an equivalent circuit. The most EMITTER MILLIAMPERES (te) Figure 8 EMITTER CHARACTERISTIC CURVE FOR TYPICAL POINT CONTACT TRANSISTOR JUNCTION ISTOR IMA VCR SV.) re -EMITTER 1O0ß SOA Cb -SASE 300A SOOA RESISTANCE Figure 7 OUTPUT CHARACTERISTICS OF TYPICAL JUNCTION TRANSISTOR (LE RESISTANCE RESCSÁL10EOR c4- CURRENT AMPLIFICATION 20000A 2.0 1 MEGONM 0.57 Figure 9 LOW FREQUENCY EQUIVALENT (Common Bose) CIRCUIT FOR POINT CONTACT AND JUNCTION TRANSISTOR convenient equivalent circuit for the low frequency small signal performance of both pointcontact and junction transistors is shown in figure 9. r., rN, and rT, are dynamic resistances which can be associated with the emitter, base and collector regions of the transistor. The current generator aI., represents the transport of charge from emitter to collector. Typical values of the equivalent circuit are shown in figure 9. Transistor Configurations There are three basic transistor configurations: grounded base connection, grounded emitter connection, and grounded collector connection. These correspond roughly to grounded grid, grounded cathode, and grounded plate circuits in vacuum tube terminology (figure 10) . The grounded base circuit has a low input impedance and high output impedance, and no phase reversal of signal from input to output circuit. The grounded emitter circuit has a higher input impedance and a lower output impedance than the grounded base circuit, and a reversal of phase between the input and output signal occurs. This circuit usually provides maximum voltage gain from a transistor. The grounded collector circuit has relatively high input impedance, low output impedance, and no phase reversal of signal from input to output circuit. Power and voltage gain are both low. Figure 11 illustrates some practical vacuum tube circuits, as applied to transistors. www.americanradiohistory.com Transistors and Semi -Conductors 96 GROUNDED EMITTER CONNECTION GROUNDED BASE CONNECTION THE RADIO GROUNDED COLLECTOR CONNECTION Figure 10 COMPARISON OF BASIC VACUUM TUBE AND TRANSISTOR CONFIGURATIONS 5 -5 Transistor Circuitry To establish the correct operating parameters of the transistor, a bias voltage must be established between the emitter and the base. Since transistors are temperature sensitive devices, and since some variation in characteristics usually exists between transistors of a given type, attention must be given to the bias system to FLIP -FLOP COUNTER R. F. overcome these difficulties. The simple self -bias system is shown in figure 12A. The base is simply connected to the power supply through a large resistance which supplies a fixed value of base current to the transistor. This bias system is extremely sensitive to the current transfer ratio of the transistor, and must be adjusted for optimum results with each transistor. When the supply voltage is fairly high and OSCILLATOR ONE -STAGE RECEIVER RFC CRYSTAL OSCILLATOR BLOCKING OSCILLATOR DIRECT -COUPLED AMPLIFIER Figure 11 TYPICAL TRANSISTOR CIRCUITS www.americanradiohistory.com AUDIO AMPLIF ER HANDBOOK Transistor Circuitry 97 -E E BIAS BIAS LOAD RESISTOR RESISTOR RESISTOR LOAD RESISTOR LOAD RESISTOR R2= lo Re Re = soo- +000 2 SO + e O n Li (REVERSE POLARITY FOR NPN TRANSISTOR) O Figure 12 BIAS CONFIGURATIONS FOR TRANSISTORS. The voltage divider system of C is recommended for general transistor use. Ratio of establishes base bias, and emitter bias is provided by voltage drop across Re. Battery Polarity is reversed for N -P -N transistors. wide variations in ambient temperature do not occur, the bias system of figure 12B may be used, with the bias resistor connected from base to collector. When the collector voltage is high, the base current is increased, moving the operating point of the transistor down the load line. If the collector voltage is low. the operating point moves upwards along the load line, thus providing automatic control of the base bias voltage. This circuit is sensitive to changes in ambient temperature, and may permit transistor failure when the transistor is operated near maximum dissipation ratings. A better bias system is shown in figure 12C, where the base bias is obtained from a voltage divider, (R1, R2 ), and the emitter is forward biased. To prevent signal degeneration, the emitter bias resistor is bypassed with a large capacitance. A high degree of circuit stability is provided by this form of bias, providing the emitter capacitance is of the order of 50 t+fd. for audio frequency applications. Audio Circuitry A simple voltage amplifier is shown in figure 13. Distabilization is employed in the rect current enitter circuit. Operating parameters for the R1 /R: amplifier are given in the drawing. In this case, the input impedance of the amplifier is quite low. When used with a high impedance driving source such as a crystal microphone a step down input transformer should be employed as shown in figure 13B. The grounded collector circuit of figure 13C provides a high input impedance and a low output impedance, much as in the manner of a vacuum tube cathode follower. The circuit of a two stage resistance coupled amplifier is shown in figure 14A. The input impedance is approximately 1100 ohms. Feedback may be placed around this amplifier from the emitter of the second stage to the base of the first stage, as shown in figure 14B. A direct coupled version of the r -c amplifier is shown in figure 14C. The input impedance is of the order of 15,000 ohms, and an overall voltage gain of 80 may be obtained with a supply potential of 12 volts. It is possible to employ N -P -N and P -N -P transistors in complementary symmetry circuits which have no equivalent in vacuum tube design. Figure 15A illustrates such a circuit. A symmetrical push -pull circuit is shown in -t2V VOLTAGE GAIN = 80 INPUT IMPEDANCE L 1200 VOLTAGE GAI N P -N . 0 97 INPUT IMPEDANCE 'L 300 /l 11 Figure 13 VOLTAGE AMPLIFIERS -P TRANSISTOR 11 A resistance coupled amplifier employing an inexpensive CK -722 transistor is shown in A. For use with a high impedance crystal microphone, a step -down transformer matches the low input impedance of the transistor, as shown in B. The grounded collector configuration of C provides an input impedance of about 300,000 ohms. www.americanradiohistory.com 98 THE RADIO Transistors and Semi -Conductors 4.7N 6A/N'L IOUF R2 Figure 14 TWO STAGE TRANSISTOR AUDIO AMPLIFIERS The feedback loop of B may be added to the r -c amplifier to reduce distortion, or to control the audio response. A direct coupled amplifier is shown in C. figure 15B. This circuit may be used to directly drive a high impedance loudspeaker, eliminating the output transformer. A direct coupled three stage amplifier having a gain figure of 80 db is shown in figure 15C. The transistor may also be used as a class A power amplifier, as shown in figure 16A. Commercial transistors are available that will provide five or six watts of audio power when operating from a 12 volt supply. The smaller units provide power levels of a few milli watts. The correct operating point is chosen so that the output signal can swing equally in the positive and negative directions, as shown in the collector curves of figure 16B. The proper primary impedance of the output transformer depends upon the amount of power to be delivered to the load: RP E: 2R. The collector current bias is: Ic-= 213" E, In a class A output stage, the maximum a -c power output obtainable is limited to 0.5 the allowable dissipation of the transistor. The product I, E, determines the maximum collector dissipation, and a plot of these values is shown in figure 16B. The load line should always lie under the dissipation curve, and should encompass the maximum possible area between the axes of the graph for maximum output condition. In general, the load line is tangent to the dissipation curve and passes through the supply voltage point at zero collector current. The d -c operating point is thus approximately one -half the supply voltage. The circuit of a typical push -pull class B transistor amplifier is shown in figure 17A. Push -pull operation is desirable for transistor operation, since the even -order harmonics are largely eliminated. This permits transistors to be driven into high collector current regions without distortion normally caused by non linearity of the collector. Cross -over distortion is reduced to a minimum by providing a slight forward base bias in addition to the normal emitter bias. The base bias is usually less than 0.5 volt in most cases. Excessive base bias will boost the quiescent collector current and thereby lower the overall efficiency of the stage. 2N78 NPN PNP NPN 2N78 NPN 2N77 PNP +E só_11 SPEAKER O O Figure 15 COMPLEMENTARY SYMMETRY AMPLIFIERS. N -P -N and P -N -P transistors may be combined in circuits which have no equivalent in vacuum tube design. Direct coupling between cascaded stages using a single power supply source may be employed, as in C. Impedance of power supply should be extremely low. www.americanradiohistory.com HANDBOOK Transistor Circuitry 99 2N 187A I MAX MAXIMUM COLLECTOR DISSIPATION (IC X EC) Figure 16 TYPICAL CLASS -A AUDIO POWER OPERATING POINT TRANSISTOR CIRCUIT. The correct operating point is :hosen so that output signal can swing equally in a positive or negative direction, without ex:ceding maximum collector dis- The operating point of the class B amplifier is set on the I. =O axis at the point where the collector voltage equals the supply voltage. The collector to collector impedance of the output transformer is: Rc-. = 2Er' Po In the class B circuit, the maximum a -c power input is approximately equal to five times the allowable collector dissipation of each transistor. Power transistors, such as the 2N301 have collector dissipation ratings of 5.5 watts and operate with class B efficiency of about 67%. To achieve this level of operation the heavy duty transistor relies upon efficient heat transfer from the transistor case to the chassis, using the large thermal capacity of the chassis as a heat sink. An infinite heat sink may be approximated by mounting the transistor in the center of a 6" x 6" copper or aluminum sheet. This area may be part of a 'arger chassis. The collector of most power transistors is electrically connected to the case. For applications where the collector is not grounded a thin sheet of mica may be used between the case of the transistor and the chassis. Power transistors such as the Philco T -1041 may be used in the common collector class B a.7N configuration (figure 17C) to obtain high power output at very low distortions comparable with those found in quality vacuum tube circuits having heavy overall feedback. In addition, the transistor may be directly bolted to the chassis, assuming a negative grounded power supply Power output is of the order of 10 watts, with about 0.5% total distortion. Circuitry Transistors may be used for radio frequency work provided the alpha cutoff frequency of the units is sufficiently higher than the operating frequency. Shown in figure 18A is a typical i -f amplifier employing an N -P -N transistor. The collector current is determined by a voltage divider on the base circuit and by a bias resistor in the emitter leg. Input and output are coupled by means of tuned i -f transformers. Bypass capacitors are placed across the bias resistors to prevent signal frequency degeneration. The base is connected to a low impedance untuned winding of the input transformer, and the collector is connected to a tap on the output transformer to provide proper matching, and also to make the performance of the stage relatively independent of variations between transistors of the same type. With a rate -grown N -P -N transistor such as the G.E. 2N293, it is unnecessary to use neutralization to obtain circuit stability. When P -N -P alloy R -F 2N225 V T-104_1 12V. ZP-soonc.T. ZS'3000CT. 2Ec Ec COLLECTOR VOLTAGE sipation. z ZS= LOAD BOO n LINE 200 MW NO SIGNAL OPERATING POINT J 2N109 -T R1 COLLECTOR VOLTAGE EC SO ADJUST Ri FOR V. BASE BIAS O.4 1 X ICC ICC í13v. - 0.3 AMP. (NAB.)',.35A. PO' 10 WATTS Figure 17 CLASS -B AUDIO AMPLIFIER CIRCUITRY. C permits the The common collector circuit of transistor to be bolted directly to the chassis for efficient heat transfer from the transistor case to the chassis. 100 Transistors and Semi- Conductors 2N293 N THE RADIO 2N135 PN p TO e I OUT. MIXER OR P. OUT. TO MIXER OR CONVERTER CONVERTER T N r9v Figure 18 TRANSISTORIZED I -F AMPLIFIERS. Typical transistor must be neutralized because of high collector capacitance. grown N -P -N transistor does not usually require external neutralizing circuit. P -N -P Rate 1N64 Figure 19 AUTOMATIC VOLUME CONTROL CIRCUIT ro MIXER OR CONVERTER FOR TRANSISTORIZED I -F AMPLIFIER. transistors are used, it is necessary to neutralize the circuit to obtain stability (figure 18B). The gain of a transistor i -f amplifier will decrease as the emitter current is decreased. This transistor property can be used to control the gain of an i -f amplifier so that weak and strong signals will produce the same audio output. A typical i -f strip incorporating this automatic volume control action is shown in figure 19. R -f transistors may be used as mixers or autodyne converters much in the same manner as vacuum tubes The autodyne circuit is shown in figure 20. Transformer T, feeds back a signal from the collector to the emitter causing oscillation. Capacitor C, tunes thé oscillator circuit to a frequency 455 kc. higher than that of the incoming signal. The local oscillator signal is inductively coupled into the emitter circuit of the transistor. The incoming signal is resonated in T_ and coupled via a low impedance winding to the base circuit. Notice that the base is biased by a voltage divider circuit much the same as is used in audio frequency operation. The two signals are mixed in this stage and the desired beat frequency of 455 kc. is selected by i -f transformer T1 and passed to the next stage. Collector currents of 0.6 ma. to 0.8 ma. are common, and the local oscillator injection voltage at the emitter is in the range of 0.15 to 0.25 volts, r.m.s. A complete receiver "front end" capable of operation up to 23 Mc. is shown in figure 21. The RCA 2N247 drift transistor is used for the r -f amplifier (TRI ), mixer (TR2), and high frequency oscillator (TR3) The 2N247 incorporates an interlead shield, cutting the interlead capacitance to .003 q fd. If proper shielding is employed between the tuned circuits of the r -f stage, no neutralization of the stage is required. The complete assembly obtains power from a 9 -volt transistor battery. Note that input and output circuits of the transistors are tapped at low impedance points on the r -f coils to achieve proper impedance match. . Figure 20 THE AUTODYNE CONVERTER CIRCUIT USING A 2N168A AS A MIXER. www.americanradiohistory.com HANDBOOK RF Transistor Circuitry 101 Figure 21 AMPLIFIER, MIXER, AND OSCILLATOR STAGES FOR TRANSISTORIZED HIGH FREQUENCY RECEIVER. THE RCA 2N247 DRIFT TRANSISTOR IS CAPABLE OF EFFICIENT OPERATION UP TO 23 Mc. i L Sufficient coupling of the proper input and output circuits of the transistor will permit oscillation up to and slightly above the alpha cutoff frequency. Various forms of transistor oscillators are shown in figure 22. A simple grounded emitter Hartley oscillator having positive feedback between the base and the collector (22A) is compared to a grounded base Hartley oscillator (22B) . In each case the resonant tank circuit is common to the input and output circuits of the transistor. Self bias of the transistor is employed in both these circuits A more sophisticated oscillator employing a 2N247 transistor and utilizing a voltage divider -type bias system (figure 22C) is capable of operation up to 50 Mc. or so. The tuned circuit is placed in the collector, with a small emitter -collector capacitor providing feedback to the emitter electrode. A P -N -P and an N -P -N transistor may be combined to form a complementary Hartley oscillator of high stability ( figure 23). The collector of the P -N -P transistor is directly Transistor Oscillators phase between coupled to the base of the N -P -N transistor, and the emitter of the N -P -N transistor furnishes the correct phase reversal to sustain oscillation. Heavy feedback is maintained between the emitter of the P -N -P transistor and the collector of N -P -N transistor. The degree of feedback is controlled by R1. The emitter resistor of the second transistor is placed at the +9V. PwP 2N247 1N81 P -N 1N81 ion NPN 2N78 Figure 23 COMPLEMENTARY HARTLEY OSCILLATOR -P and N -P -N transistors bility form high sta- oscillator. Feedback between P -N -P emitter and N -P -N collector is controlled by R,. 1N81 diodes are used as amplitude limiters. Frequency of oscillation is determined by L, C, -C :. RFC RFC Figure 22 TYPICAL TRANSISTOR OSCILLATOR CIRCUITS A- Grounded Emitter Hartley Grounded Base Hartley C -2N247 Oscillator Suitable for B- SO Mc. operation. www.americanradiohistory.com 102 THE RADIO Transistors and Semi -Conductors 2N33 POiNT-CONTA.T TRANS.STON POINT -CONTACT TRANSISTOR LE CI.ARGN PER OO RFC :E V. Figure 25 RELAXATION OSCILLATOR USING POINT- CONTACT OR SURFACE Figure 24 NEGATIVE RESISTANCE OF POINT -CONTACT TRANSISTOR PERMITS HIGH FREQUENCY OSCILLATION (50 Mc) WITHOUT WITHOUT NECESSITY OF EXTERNAL FEEDBACK PATH. BARRIER TRANSISTORS. Relaxation Oscillators Transistors have almost unlimited use in relaxation acid R -C oscillator service. The negative re- sistance characteristic of the point contact transistor make it well suited to such application. Surface barrier transistors are also widely used in this service, as they have the highest alpha cutoff frequency among the group of -alphaless-than-unity- transistors. Relaxation oscillators used for high speed counting require transistors capable of operation at repetition rates of 5 Mc. to 10 Mc. center of the oscillator coil to eliminate loading of the tuned circuit. Two germanium diodes are employed as amplitude limiters, further stabilizing amplifier operation. Because of the low circuit impedances, it is permissible to use extremely high -C in the oscillator tank circuit, effectively limiting oscillator temperature stability to variations in the tank inductance. The point- contact transistor exhibits negative input and output resistances over part of its operaing range, due to its unique ability to multiply the input current. This characteristic affords the use of oscillator circuitry having no external feedback paths ( figure 24). A high impedance resonant circuit in the base lead produces circuit instability and oscillation at the resonant frequency of the L -C circuit. Positive emitter bias is used to insure thermal circuit stability. A simple emitter controlled relaxation oscillator is shown in figure 25, together with its operating characteristic. The emitter of the transistor is biased to cutoff at the start of the cycle (point I) The charge on the emitter capacitor slowly leaks to ground through the emitter resistor, R1. Discharge time is determined by the time constant of RICI. When the emitter voltage drops sufficiently low to permit the transistor to reach the negative resistance region (point 2) the emitter and collector resistances drop to a low value, and the collector . *E e. PO5ITiwE TRi.,,ER PJLSE EPUL 5E M OUT NPN - 1/111 NP P N P lON Figure 26 TRANSISTORIZED BLOCKING OSCILLATOR (A) AND ECCLES -JORDAN BI- STABLE MULTIVIBRATOR (B). High -alpha transistors must be employed in counting circuits to reduce effects of storage time caused by transit lag in transistor base. www.americanradiohistory.com HANDBOOK Transistor Circuitry 103 2n PHONE`, C1- 123LLF, .W. M /LLER # 2110 C2- 791/11F, PART OF C I L1 - "LOOPSTIC K. COIL, J.W MILLER La- OSCILLATOR COIL, J W. I Figure 28 SCHEMATIC, TRANSISTORIZED BROADCAST BAND (S00 DYNE RECEIVER. " L OOP ST ICI COIL Figure 27 "WRIST RADIO" CAN BE MADE WITH LOOPSTICK, DIODE, AND INEXPENSIVE CK -722 TRANSISTOR. A TWENTY FOOT ANTENNA WIRE WILL PROVIDE GOOD RECEPTION IN STRONG SIGNAL AREAS. current is limited only by the collector resistor, R. The collector current is abruptly reduced by the charging action of the emitter capacitor CI (point 3), bringing the circuit back to the original operating point. The "spike" of collector current is produced during the charging period of C. The duration of the pulse and the pulse repetition frequency (p.r.f.) are controlled by the values of C, R1, R_, and R. Transistors may also be used as blocking oscillators (figure 26A) . The oscillator may be synchronized by coupling the locking signal to the base circuit of the transistor. An oscillator of this type may be used to drive a flip flop circuit as a counter. An Eccles -Jordan bi- stable flip -flop circuit employing surface barrier transistors may be driven between "off" and "on" positions by an exciting pulse as shown in figure 26B. The first pulse drives the "on" transistor into saturation. This transistor remains in a highly conductive state until the second exciting pulse arrives. The transistor does not immediately return to the cut -off - #2003 MILLER 4 2002 TI -4SS RC. I.F. TRANSFORMER, T2 -455 PI C. F. TRANSFORMER, J.W MILLER2031 J.W. MILLER 2032 1600 KC.) SUPERHETERO- state, since a time lapse occurs before the output waveform starts to decrease. This storage time is caused by the transit lag of the minority carriers in the base of the transistor. Proper circuit design and the use of high -alpha transistors can reduce the effects of storage time to a minimum. Driving pulses may be coupled to the multivibrator through steering diodes as shown in the illustration. 5 -6 Transistor Circuits With the introduction of the dollar transistor, many interesting and unusual experiments and circuits may be built up by the beginner in the transistor field. One of the most interesting is the "wrist watch" receiver, illustrated in figure 27. A diode and a transistor amplifier form a miniature broadcast receiver, which may be built in a small box and carried on the person. A single 1.5 -volt penlite cell provides power for the transistor, and a short length of antenna wire will suffice in the vicinity of a local broadcasting station. A transistorized superhetrodyne for broadcast reception is shown in figure 28. No antenna is required, as a ferrite "loop-stick" is used for the r -f input circuit of the 2N136 mixer transistor. A miniature magnetic "hearing aid" type earphone may be employed with this receiver. A simple phonograph amplifier designed for use with a high impedance crystal pickup is shown in figure 29. Two stages of amplification using 2N109 transistors are used to drive two 2N109 transistors in a class B con- figuration. Approximately 200 milliwatts of www.americanradiohistory.com 104 Transistors and Semi -Conductors 2N109 2N1O9 220N CRYSTAL PICKUP f0 0 i5M 2UF 2N109 0 27M K Figure 29 HIGH GAIN, LOW DISTORTION AUDIO AMPLIFIER, SUITABLE FOR USE WITH A CRYSTAL PICKUP. POWER OUTPUT IS 250 MILLIWATTS. power may be obtained with a battery supply of 12 volts. Peak current drain under maximum signal conditions is 40 ma. Shown in figure 30 is an inexpensive and compact 25 watt transistorized modulator suitable for mobile use with an automobile having a 12 volt ignition system. This unit may be used to modulate a 6146 r.f. amplifier stage running at 400 volts and 125 milliamperes plate power input. The two DS -501 power transistors (Delco) are mounted on a heat sink made of a 6" x 6" x 1/8" aluminum plate. The components are mounted on the sink which serves as the chassis. Output transformer T1 -I50 T., consists of a 6.3 volt filament transformer with the ends of the low voltage winding connected to the collectors of the output transistors. Resting modulator current is about 0.7 amperes, rising to nearly 2 amperes on full modulation peaks. The modulator should be positioned so that motor heat and warm air is deflected from the unit, or the efficiency of the aluminum heat sink will be impaired and damage to the output transistors may result. A good location for the unit is under the dash against the firewall. Microphone gain may be adjusted by changing the value of the 100 ohm, 2 watt series resistor. Figure 30. TRANSISTORIZED 25 WATT MOBILE MODULATOR. ohm primary, 490 ohm secondary (center tap on primary not used), Thordarson TR -S. T2-400 ohm primary, 4 and 16 ohm secondary. Stancor TA -41. T3-6.3 volt center tap, 3a. (See text.) Note-Output transistors are insulated from heat-sink by Delco (s: 1221264). insulators mica R.F. LOAD NOTE: C /ACV /T RETURNS ro BATTERY TERMINAL. CONTROL -Nor- CONTROL CIRCUIT +12.e V. www.americanradiohistory.com -1z.e V. Zener Diodes 5 -7 Zener Diodes The Zener Diode is a semiconductor device that can be used as a constant voltage reference, or as a control element. Zener diodes are available in ratings to 50 watts, with zener voltages of approximately 4 volts to 200 volts. The zener diode has electrical characteristics that are derived from a rectifying junction which operates at a reverse bias condition not normally used. The zener knee (figure 31) and constant voltage plateau are obtained when this rectifying junction is back -biased above the junction breakdown voltage. The break from non -conductance to conductance is very sharp. At applied voltages greater than the breakdown point, the voltage drop across the diode junction becomes essentially constant for a relatively wide range of currents. This is the zener control region. Thermal dissipation is obtained by mounting the zener diode to a heat sink composed of a large area of metal having free access to ambient air. - ZENER KNEE MA. CONSTANT VOLTAGE Mi M _ CURRENT 1.S 1S 2 VP (VOLTS) u REVERSe CHARACTERISTIC S= {M AM. N FIGURE 31 BETWEEN .ZENER KNEE- AND POINT OF MAXIMUM ZENER CUR RENT, THE ZENER VOLTAGE IS ESSENTIALLY CONSTANT AT SOVOLTS. DIODE 2 VOLT REG. VOLT. FIGURE 32 -ZENER DIODE FUNCTIONS AS VOLTAGE REGULATOR OVER RANGE OF CONSTANT VOLTAGE PLATEAU. B -TWO ZENER DIODES OF DIFFER OS 1.0 MAX.ZENER 10 v 1LOUT MM iM= MlAT 5 The zener diode may be employed as a shunt regulator (figure 32A) in the manner of a typical "VR- tube." Two zener diodes may be employed in the circuit of figure 32B to supply very low values of regulated voltage. Two opposed zener diodes can be used to provide a.c. clipping of both halves of the cycle (figure 32C) . Zener diodes may also be used to protect meter movements as they provide a very low resistance shunt across the movement when the applied voltage exceeds a certain critical level. IN ti AMMO REVERSE VOLTAGE 10 30 20 Applications A 2 10 CHARACTERISTIC Zener Diode UNREG. VOLT qM/iI MMMMM ,. MUM= 105 - ENT VOLTAGE CAN PROVIDE SMALL REGULATED VOLTAGE. C- OPPOSED ZENER DIODES CLIP BOTH HALVES OF CYCLE OF A.C. WAVE. www.americanradiohistory.com CHAPTER SIX Vacuum Tube Amplifiers 6 -1 Vacuum Tube Parameters Electrode Potentials Symbols for Vacuum -Tube Parameters E« ep eg Ep Eg grid- screen conversion `Bp Cpi, Cm Cou, I, fpm ipmax igmaz Ip Ig factor tube) grid grid voltage cutoff average grid current fundamental maximum maximum grid current Other Symbols grid- cathode grid Pi plate- cathode input output grid minimum maximum -- average plate current -peak plate -instantaneous plate current instantaneous -- static plate current static - plate P.-plate -- plate dissipation plus bias losses) lb Intere!ectrode Capacitances C¡gk -- instantaneous plate potential instantaneous potential instantaneous plate voltage -positive instantaneous voltage static plate voltage ---static bias Electrode Currents factor mu - eco Gm Gc grid supply voltage (a negative epmin egmp Tube Constants --transconductance plate resistance /tu transconductance(mixer -- -plate capacitance capacitance capacitance --- capacitance (tetrode capacitance (tetrode -d-c peak grid excitation voltage (1/2 total peak -to -peak grid swing) Epm -peak plate voltage (! i total peak -to -peak plate swing) Egm involving vacuum -tube parameters, the following symbols will be used throughout this book: R, plate supply voltage (a positive quantity) quantity) As an assistance in simplify ing and shortening expressions ft- amplification -d-c Ebb The ability of the control grid of a vacuum tube to control large amounts of plate power with a small amount of grid energy allows the vacuum tube to be used as an amplifier. It is this ability of vacuum tube s to amplify an extremely small amount of energy up to almost any level without change in anything except amplitude which makes the vacuum tube such an extremely valuable adjunct to modern electronics and communication. or pentode) or pentode) Pp Pd power input power output grid driving power (grid 106 www.americanradiohistory.com current grid current Classes of Amplifiers ST -F-_, r-- r--1 17-T- Cw --- Ccv:7: - I .7.-.7. CcKi_ :=. L Clan CiN :t: -,Cour I __ _J 1 PENTODE OR TETRODE TRIODE Figure STATIC INTERELECTRODE TANCES WITHIN A TRIODE, OR TETRODE 1 CAPACIPENTODE, 107 is the grid -to -plate capacitance, and A is the stage gain. This expression assumes that the vacuum tube is operating into a resistive load such as would be the case with an audio stage working into a resistance plate load in the middle audio range. The more complete expression for the input admittance (vector sum of capacitance and resistance) of an amplifier operating into any type of plate load is as follows: Cgp Input capacitance = Cgk + (1 + A cos (9) Cgp Input resistance 135-grid dissipation Np 9p 9g R1 Z1 --load - plate efficiency(expressed as a decimal) one -half angle of plate current flow one -half angle of grid current flow resistance load impedance A The relationships between certain of the electrode potentials and currents within a vacuum tube are reasonably constant under specified alone angle of the plate load impedance, positive for inductive loads, negative for capacitive Vacuum-Tube conditions of operation. These relationships are called vacuum -tube constants and are listed in the data published by the manufacturers of vacuum tubes. The defining equations for the basic vacuum -tube constants are given in Chapter Four. Interelectrode The values of interelectrode Capacitances and capacitance published in Miller Effect vacuum -tube tables are the static values measured, in the case of triodes for example, as shown in figure 1. The static capacitances are simply as shown in the drawing, but when a tube is operating as amplifier there is another consideration known as Miller Effect which causes the dynamic input capacitance to be different from the static value. The output capacitance of an amplifier is essentially the same as the static value given in the published tube tables. The grid -to-plate capacitance is also the same as the published static value, but since the Cgp acts as a small capacitance coupling energy back from the pl ate circuit to the grid circuit, the dynamic input capacitance is equal ro the static value plus an amount (frequently much greater in the case of a triode) determined by the gain of the stage, the plate load impedance, and the Cgp feedback capacitance. The total value for an audio amplifier stage can be expressed in the following equation: (dynamic) _ ( static) + (A + 1) Cgp where Co is the grid -to- cathode capacitance, O = phase O Constants sin Where: Cgk = grid -to- cathode capacitance Cgp = grid -to-plate capacitance A = voltage amplification of the tube It can be seen from the above that if the plate load impedance of the stage is capacitive or inductive, there will be a resistive component in the input admittance of the stage. The resistive component of the input admittance will be positive (tending to load the circuit feeding the grid) if the load impedance of the plate is capacitive, or it will be negative (tending to make the stage oscillate) if the load impedance of the plate is inductive. Neutralization of Interelectrode Capacitance Neutralization of the effects of interelectrode capacitance is employed most frequently in the case of radio frequency power amplifiers. Before the introduction of the tetrode and pentode tube, triodes were employed as neutralized Class A amplifiers in receivers. This practice has been largely superseded in the present state of the art through the use of tetrode and pentode tubes in which the Cge or feedback capacitance has been reduced to such a low value that neutralization of its effects is not necessary to prevent oscillation and instability. 6 -2 Classes and Types of Vacuum -Tube Amplifiers Vacuum -tube amplifiers are grouped into various classes and sub -classes according to the type of work they are intended to perform. The difference between the various classes is determined primarily by the value of average grid bias employed and the maximum value of www.americanradiohistory.com 108 the grid. Vacuum Tube Amplifiers THE exciting signal to he impressed upon the A Class A amplifier is an amplifier biased and supplied with excitation Class A Amplifier of such amplitude that plate current flows continuously (360° of the exciting voltage waveshape) and grid current does not flow at any time. Such an amplifier is normally operated in the center of the grid- voltage plate- current transfer characteristic and gives an output waveshape which is a substantial replica of the input waveshape. Class Al Amplifier This is another term applied to the Class A amplifier in which grid current does not flow over any portion of the input wave cycle. This is a Class A amplifier operated under such conditions that the grid is driven positive over a portion of the input voltage cycle, but plate current still flows over the entire cycle. Class A2 Amplifier Class AB1 This is an amplifier operated under Amplifier such conditions of grid bias and exciting voltage that plate current flows for more than one-half the input voltage cycle but for less than the complete cycle. In other words the operating angle of plate current flow is appreciably greater than 180° but less than 360°. The suffix 1 indicates that grid current does not flow over any portion of the input cycle. Class AB2 amplifier is operated under essentially the same conditions of grid bias as the Class AB t amplifier mentioned above, but the exciting voltage is of such amplitude that grid current flows over an appreciable portion of the input wave cycle. Class AB2 A Amplifier amplifier is biased sub stantially to cutoff of plate current (without exciting voltage) so that plate current flows essentially over one -half Class B A Class B Amplifier the input voltage cycle. The operating angle of plate current flow is essentially 180 °. The Class B amplifier is almost always excited to such an extent that grid current flows. Class C amplifier is biased to a Amplifier value greater than the value required for plate current cutoff and is excited with a signal of such amplitude that grid current flows over an appreciable period of the input voltage waveshape. The angle of plate current flow in a Class C amplifier is appreciably less than 180 °, or in other words, plate current flows appreciably Class RADIO C A Figure 2 TYPES OF BIAS SYSTEMS A B - C - Grid bias Cathode bias Grid leak bias less than one -half the time. Actually, the conventional operating conditions for a Class C amplifier are such that plate current flows for 120° to 150° of the exciting voltage waveshape. There are three general types of amplifier circuits in use. These types are classified on the basis of the return for the input and output circuits. Conventional amplifiers are called cathode return amplifiers since the cathode is effectively grounded and acts as the common return for both the input and output circuits. The second type is known as a plate return amplifier or cathode follower since the plate circuit is effectively at ground for the input and output signal voltages and the output voltage or power is taken between cathode and plate. The third type is called a grid -return or grounded grid amplifier since the grid is effectively at ground potential for input and output signals and output is taken between grid and plate. Types of Amplifiers 6 -3 Biasing Methods The difference of potential between grid and cathode is called the grid bias of a vacuum tube. There are three general methods of providing this bias voltage. In each of these methods the purpose is to establish the grid at a potential with respect to the cathode which will place the tube in the desired operating condition as determined by its charac- teristics. Grid bias may be obtained from a source of voltage especially provided for this purpose, as a battery or other d -c power supply. This method is illustrated in figure 2A, and is known as fixed bias. A second biasing method is illustrated in figure 2B which utilizes a cathode resistor across which an IR drop is developed as a result of plate current flowing through it. The www.americanradiohistory.com Amplifier HANDBOOK cathode of the tube is held at a positive potential with respect to ground by the amount of the IR drop because the grid is at ground potential. Since the biasing voltage depends upon the flow of plate current the tube cannot be held in a cutoff condition by means of the cat bode bias voltage developed across the cathode resistor. The value of this resistor is determined by the bias required and the plate current which flows at this value of bias, as found from the tube characteristic curves. A capacitor is shunted across the bias resistor to provide a low impedance path to ground for the a -c component of the plate current which results from an a-c input signal on the grid. The third method of providing a biasing voltage is shown in figure 2C, and is called grid -leak bias. During the portion of the input cycle which causes the grid to be positive with respect to the cathode, grid current flows from cathode to grid, charging capacitor C,. When the grid draws current, the grid -to- cathode resistance of the tube drops from an infinite value to a very low value, on the order of 1,000 ohms or so, making the charging time constant of the capacitor very short. This enables Cs to charge up to essentially the full value of the positive input voltage and results in the grid (which is connected to the low potential plate of the capacitor) being held essentially at ground potential. During the negative swing of the input signal no grid current flows and the discharge path of Cg is through the grid resistance which has a value of 500,000 ohms or so. The discharge time constant for C5 is, therefore, very long in comparison to the period of the input signal and only a small part of the charge on C5 is lost. Thus, the bias voltage developed by the discharge of Cs is substantially constant and the grid is not permitted to follow the positive portions of the input signal. Distortion in Amplifiers 6 -4 There are three main types of distortion that may occur in amplifiers: frequency distortion, phase distortion and amplitude distortion. distortion may occur when some frequency components of a signal are amplified more than Frequency distortion occurs at low Frequency Distortion Distortiob 109 OUTPUT SIGNAL Figure 3 Illustration of the effect of phase distortion on input wave containing o third harmonic signal two stage amplifier. Although the amplitudes of both components are amplified by identical ratios, the output waveshape is considerably different from the input signal because the phase of the third harmonic signal has been shifted with respect to the fundamental signal. This phase shift is known as phase distortion, and is caused principally by the coupling circuits between the stages of the amplifier. Most coupling circuits shift the phase of a sine wave, but this has no effect on the shape of the output wave. However, when a complex wave is passed through the same coupling circuit, each component frequency of the waveshape may be shifted in phase by a different amount so that the output wave is not a faithful reproduction of the input waveshape. a Amplitude Distortion If a signal is passed through a vac uum tube that is operating on any non -linear part of its characteristic, amplitude distortion will occur. In such a region, a change in grid voltage does not result in a change in plate current which is directly proportional to the change in grid voltage. For example, if an amplifies is excited with a signal that overdrives the tubes, the resultant signal is distorted in amplitude, since the tubes operate over a non -linear portion of their characteristic. Frequency others. frequencies if coupling capacitors between stages are too small, or may occur at high frequencies as a result of the shunting effects of the distributed capacities in the circuit. Phase Distortion input signal con sisting of a fundamental and a third harmonic is passed through In figure 3 an 6 -5 Resistance Capacitance Coupled Audio -Frequency Amplifiers Present practice in the design of audio-frequency voltage amplifiers is almost exclusively to use resistance -capacitance coupling between the low -level stages. Both triodes and www.americanradiohistory.com 1 1 Vacuum 0 Tube A mp l i fie T H E r s Figure 4 CIRCUIT FOR RESISTANCE CAPACITANCE COUPLED TRIODE AMSTANDARD PLIFIER STAGE R A D The voltage gain per stage of a resistance -capacitance coupled triode amplifier can be calculated with the aid of the equivalent circuits and expressions for the mid -frequency, high frequency, and low- frequency range given in figure 5. A triode R -C coupled amplifier stage is normally operated with values of cathode resistor and plate load resistor such that the actual voltage on the tube is approximately one -half the d -c plate supply voltage. To per Stage will be are used; triode amplifier stages discussed first. R -C Coupled Triode Stages 4 illustrates the stand circuit for a resistance - Figure and capacitance coupled amplifier stage utilizing a triode tube with cathode bias. In conventional audio-frequency amplifier design such stages are used at medium voltage G E A_ A) RP RL RG (RL+RC)+RL RG 11EG MID FREQUENCY RANGE CGN (DYNAMIC, NEXT STAGE) L=-LEG - A HIGH FREE). A MID FREE). 1 Ni 1+ (REQ /XS)2 RL R CO 1+ HIGH FREQUENCY RANGE Xs G E= -L O levels (from 0.01 to 5 volts peak on the grid of the tube) and use medium -p triodes such as the 6J5 or high -p triodes such as the 6SF5 or 6SL7 -GT. Normal voltage gain for a single stage of this type is from 10 to 70, depending upon the tube chosen and its operating conditions. Triode tubes are normally used in the last voltage amplifier stage of an R -C amplifier since their harmonic distortion with large output voltage (25 to 75 volts) is less than with a pentode tube. Voltage Gain pentodes I A LOW FREQ. A MID FREQ. ' RL RL Rn RG 2TTF (CPN+CGN (orNAMlc) = 1+ (XC /R)2 EG Xc - R = RG+ 1 2 TTFCC LOW FREQUENCY RANGE RL RP RL+ RP Figure 5 Equivalent circuits and gain equations for a triode R -C coupled amplifier stage. In using these equations, be sure to select the values of mu and RP which are proper for the static current and voltages with which the tube will operate. These values may be obtained from curves published in the RCA Tube Handbook RC -16. www.americanradiohistory.com '7IANDBOOK R -C Amplifiers Coupled 111 such as the 6SJ7. Normal voltage gain for a stage of this type is from 60 to 250, depending upon the tube chosen and its operating conditions. Pentode tubes are ordinarily used the first stage of an R -C amplifier where the high gain which they afford is of greatest advantage and where only a small voltage output is required from the stage. Figure 6 CIRCUIT FOR RESISTANCE CAPACITANCE COUPLED PENTODE AMSTANDARD PLIFIER STAGE assist the designer of such stages, data on operating conditions for commonly used tubes is published in the RCA Tube Handbook RC -16. It is assumed, in the case of the gain equations of figure 5, that the cathode by -pass capacitor, Ck, has a reactance that is low with respect to the cathode resistor at the lowest frequency to be passed by the amplifier stage. Coupled Pentode Stages R -C 6 illustrates the stand circuit for a resistance - Figure and capacitance coupled pentode amplifier stage. Cathode bias is used and the screen voltage is supplied through a dropping resistor from the plate voltage supply. In conventional audio -frequency amplifier design such stages are normally used at low voltage levels (from 0.00001 to 0.1 volts peak on the grid of the tube) and use moderate -Gm pentodes The voltage gain per stage of a resistance capacitance coupled pentode amplifier can be calculated with the aid of the equivalent circuits and expressions for the mid -frequency, high- frequency, and low- frequency range given in figure 7. To assist the designer of such stages, data on operating conditions for commonly used types of tubes is published in the RCA Tube Handbook RC -16. It is assumed, in the case of the gain equations of figure 7, that the cathode by -pass capacitor, Ck, has a reactance that is low with respect to the cathode resistor at the lowest frequency to be passed by the stage. It is additionally assumed that the reactance of the screen by -pass capacitor Cd, is low with respect to the screen dropping resistor, Rd, at the lowest frequency to be passed by the amplifier stage. Cascade Voltage Amplifier Stages When voltage amplifier stages are operated in such a manner that the output voltage of the first is fed to the grid of the second, and so forth, such stages are said to be cascaded. The total voltage gain of cascaded amplifier t= -GMEc c A = GM REO RL REO Figure 7 Equivalent circuits and gain equations for a pentode R -C coupled amplifier stage. In using these equations be sure to select the values of Gm and Rp which are proper for the static currents and voltages with which the tube will operate. These values may be obtained from curves published in the RCA Tube Re RG MID FREQUENCY RANGE I= -GMEc A HIGH FRED. AMID FOtO. R CO - 1+ HIGH FREQUENCY RANGE Xs' }(REO /Xs)2 RL 1 Rc+RP 277F (CPK+CGK (DYNAMIC) A LOW FREQ. _ Handbook RC -16. A MID LOW FREQUENCY RANGE Xc R = www.americanradiohistory.com I+(XCF R)2 FOCO. 277r CC RO + RL RP RL+RP 112 Vacuum Amplifiers Tube 100- 1. THE RADIO 500000 ONMs RL 2-RL= 100000OHMS 3. RL= 4. RL' so 000 oHMs 20000 01-1M3 z30 á u 1000 100 10000 100000 MI0-EREQUENCr GAIN = GMV, RL NIGN- FREOUCNCY GAIN a Gm*, ¿COUPLING e CouT 1004000 C FREQUENCY (C.P.S V,.CINr2 FOR COMPROMISE HIGH REOOCNCV Figure XLL - 8 The variation of stage gain with frequency in an r-c coupled pentode amplifier for various values of plate load resistance WHERE Amplifier typical frequency response curve for an R -C coupled audio amplifier is shown in figure 8. It is seen that the amplification is poor for the extreme high and low frequencies. The reduced gain at the low frequencies is caused by the loss of voltage across the coupling capacitor. In some cases, a low value of coupling capacitor is deliberately chosen to reduce the response of the stage to hum, or to attenuate the lower voice frequencies for communication purposes. For high fidelity work the product of the grid resistor in ohms times the coupling capacitor in microfarads should equal 25,000. (ie.: 500,000 ohms x 0.05 µfd = 25,000). The amplification of high frequencies falls off because of the Miller effect of the subsequent stage, and the shunting effect of residual circuit capacities. Both of these effects may be minimized by the use of a low value of plate load resistor. A Response The correct operating bias for a high -mu triode such as the GSL7, is fairly critical, and will be found to be highly variable from tube to tube because of minute variations in contact potential within the tube itself. A satisfactory bias method is to use grid leak bias, with a grid resistor of one to ten meg- Grid Leak Bias for High Mu Triodes EQUALIZATION XC AT fC . XC AT f e CUTOFF FRCQUENC, OF AMPLIFIER C fC LL I PEAKING INDUCTOR POR COMPROMISE LOW PRCOUENC EQUALISATION stages is obtained by taking the product of the voltage gains of each of the successive stages. R -C S RL Re' Sometimes the voltage gain of an amplifier stage is rated in decibels. Voltage gain is converted into decibels gain through the use of the following expression: db = 20 log A, where A is the voltage gain of the stage. The total gain of cascaded voltage amplifier stages can be obtained by adding the number of decibels gain in each of the cascaded stages. 0 NETWORK T C DISTRIBUTED Ro (Goo vi RL) Rs Ce °RACK Co C a = 25 TO SO Of0 IN PARALLEL WITH D01 MICA CAPACITANCE FROM AsOAC WITH 001 MICA IN PARALLEL Figure 9 SIMPLE COMPENSATED VIDEO AMPLIFIER CIRCUIT Resistor RL in conjunction with coil LL serves to flatten the high -frequency response of the stage, while CB and R serve to equalize the low- frequency response of this simple video amplifier stage. ohms connected directly between grid and cathode of the tube. The cathode is grounded. Grid current flows at all times, and the effective input resistance is about one -half the resistance value of the grid leak. This circuit is particularly well suited as a high gain amplifier following low output devices, such as crystal microphones, or dynamic micro- phones. resistance- capacity coupled amplifier can be designed to provide a good frequency response for almost any desired range. For instance, such an amplifier can be built to provide a fairly uniform amplification for frequencies in the audio range of about 100 to 20,000 cycles. Changes in the values of coupling capacitors and load resistors can extend this frequency range to cover the very wide range required for video service. However, extension of the range can only be obtained at the cost of reduced overall amplification. Thus the R -C method of coupling allows good frequency response with minimum distortion, but low amplification. Phase distortion is less with R -C coupling R -C Amplifier General Characteristics www.americanradiohistory.com A HANDBOOK Video Frequency Amplifiers than with other types, except direct coupling. The R -C amplifier may exhibit tendencies to "motorboat" or oscillate if it is used with a high impedance plate supply. resistance -capacitance coupling is most commonly used, there are certain circuit conditions wherein coupling methods other than resistance capacitance are more effective. Transformer coupling, as illustrated in figure 1013, is seldom used at the present time between two successive single -ended stages of an audio amplifier. There are several reasons why resistance coupling is favored over transformer coupling between two successive single -ended stages. These are: (1) a transformer having frequency characteristics comparable with a properly designed R -C stage is very expensive; (2) transformers, unless they are very well shielded, will pick up inductive hum from nearby power and filament transformers; (3) the phase characteristics of step -up interstage transformers are poor, making very difficult the inclusion of a transformer of this type within a feedback loop; and (4) transformers are heavy. However, there is one circuit application where a step-up interstage transformer is of considerable assistance to the designer; this is the case where it is desired to obtain a large amount of voltage to excite the grid of a cathode follower or of a high -power Class A amplifier from a tube operating at a moderate plate voltage. Under these conditions it is possible to obtain apeak voltage on the secondary of the transformer of a value somewhat greater than the d-c plate supply voltage of the tube supplying the primary of the transformer. Transformer Coupling Video -Frequency 6 -6 Amplifiers A video -frequency amplifier is one which has been designed to pass frequencies from the lower audio range (lower limit perhaps 50 cycles) to the middle r -f range (upper limit perhaps 4 to 6 megacycles). Such amplifiers, in addition to passing such an extremely wide frequency range, must be capable of amplifying this range with a minimum of amplitude, phase, and frequency distortion. Video amplifiers are commonly used in television, pulse communication, and radar work. Tubes used in video amplifiers must have high ratio of Gm to capacitance if a usable gain per stage is to be obtained. Commonly available tubes which have been designed for or are suitable for use in video amplifiers are: 6AU6, 6AG5, 6AK5, 6CB6, 6AC7, 6AG7, and 6K6 -GT. Since, at the upper frequency limits of a video amplifier the input and output shunting capacitances of the amplifier tubes have rather low values of reactance, low values of coupling resistance along with peaking coils or other special interstage coupling impedances are usually used to flatten out the gain /frequency and hence the phase/ frequency characteristic of the amplifier. Recommended operating conditions along with expressions for calculation of gain and circuit values are given in figure 9. Only a simple two -terminal interstage coupling network is shown in this figure. The performance and gain -per -stage of a video amplifier can be improved by the use of increasingly complex two-terminal inter stage coupling networks or through the use of four -terminal coupling networks or filters between successive stages. The reader is referred to Terman's "Radio Engineer's Handbook" for design data on such interstage coupling networks. a Push -Pull Transformer transformer coupling between two stages is illustrated in figure 10C. This interstage coupling arrangement is fairly commonly used. The system is particularly effective when it is desired, as in the system just described, to obtain a fairly high voltage to excite the grids of a high power audio stage. The arrangement is also very good when it is desired to apply feedback to the grids of the push -pull stage by applying the feedback voltage to the lowpotential sides of the two push -pull secondaries. Impedance coupling between two stages is shown in figure 10D. This circuit arrangement is seldom used, but it offers one strong advantage over R -C interstage coupling. This advantage is the fact that, since the operating voltage on the tube with the impedance in the plate circuit is the plate supply voltage, it is possible to obtain approximately twice the peak voltage output that it is possible to obtain with R -C coupling. This is because, as has been Impedance Other Interstage Coupling Methods Figure 10 illustrates, in addition to resistance- capacitance interstage coupling, seven additional methods in which coupling between two successive stages of an audio -frequency amplifier may be accomplished. Although Push -pull Interstage Coupling Coupling 6 -7 113 www.americanradiohistory.com 114 Vacuum Tube Amplifiers THE RADIO +e pA RESISTANCE- CAPACITANCE COUPLING © TRANSFORMER COUPLING © PUSH -PULL TRANSFORMER COUPLING © IMPEDANCE COUPLING IMPEDANCE -TRANSFORMER COUPLING 0 CATHODE COUPLING pH +5 E© RESISTANCE- TRANSFORMER COUPLING +5 © INTERSTAGE DIR- ECT Figure 10 COUPLING METHODS FOR AUDIO FREQUENCY VOLTAGE mentioned before, the d -c plate voltage on an R -C stage is approximately one -half the plate supply voltage. These two circuit arrangements, illustrated former Coupling in figures 10E and 10F, are employed when it is desired to use transformer coupling for the reasons cited above, but where it is desired that the d -c plate current of the amplifier Impedance -Transformer and Resistance -Trans- +5 COUPLING AMPLIFIERS stage be isolated from the primary of the coupling transformer. With most types of high permeability wide -response transformers it is necessary that there be no direct -current flow through the windings of the transformer. The impedance- transformer arrangement of figure 10E will give a higher voltage output from the stage but is not often used since the plate coupling impedance (choke) must have very high inductance and very low distributed capacitance in order not to restrict the range of www.americanradiohistory.com HANDBOOK Phase Inverters 115 same type tubes with the values of plate voltage and load resistance to be used for the Gw = - GM G= RK GM 2G+1 RR RP' = RP = GM G+11 L G RP G+1 = = = (1+ ) CATHODE RESISTOR GM OF EACH TUBE Al OF EACH TUBE RP OF EACH TUBE EQUIVALENT FACTORS INDICATED ABOVE BY (I) ARE THOSE OBTAINED BY USING AN AMPLIFIER WITH A PAIR OF SIMILAR TUBE TYPES IN CIRCUIT SHOWN ABOVE. Figure 11 Equivalent factors for a pair of similar triodes operating as a cathode-coupled audio frequency voltage amplifier. the transformer which it and its associated tube feed. The resistance -transformer arrange- ordinarily quite satisfactory where it is desired to feed a transformer from a voltage amplifier stage with no d.c.in the transformer primary. ment of figure 10F is The cathode coupling arrangement of figure 10G has been widely used only comparatively recently. One outstanding characteristic of such a circuit is that there is no phase reversal between the grid and the plate circuit. All other common types of interstage coupling are accompanied by a 180° phase reversal between the grid circuit and the plate circuit of the tube. Figure 11 gives the expressions for determining the appropriate factors for an equivalent triode obtained through the use of a pair of similar triodes connected in the cathode coupled circuit shown. With these equivalent triode factors it is possible to use the expressions shown in figure 5 to determine the gain of the stage at different frequencies. The input capacitance of such a stage is less than that of one of the triodes, the effective grid to -plate capacitance is very much less (it is so much less that such a stage may be used as an r -f amplifier without neutralization), and the output capacitance is approximately equal to the grid -to -plate capacitance of one of the triode sections. This circuit is particularly effective with tubes such as the 6J6, 6N7, and 6SN7 -GT which have two similar triodes in one envelope. An appropriate value of cathode resistor to use for such a stage is the value which would be used for the cathode resistor cf a conventional amplifier using one of the Cathode Coupling cathode -coupled stage. Inspection of the equations in figure 11 shows that as the cathode resistor is made smaller, to approach zero, the Gm approaches zero, the plate resistance approaches the Rp of one tube, and the mu approaches zero. As the cathode resistor is made very large the Gm approaches one half that of a single tube of the same type, the plate resistance approaches twice that of one tube, and the mu approaches the same value as one tube. But since the Gm of each tube decreases as the cathode resistor is made larger (since the plate current will decrease on each tube) the optimum value of cathode resistor will be found to be in the vicinity of the value mentioned in the previous paragraph. Direct coupling between successive amplifier stages (plate of first stage connected directly to the grid of the succeeding stage) is complicated by the fact that the grid of an amplifier stage must be operated at an average negative potential with respect to the cathode of that stage. However, if the cathode of the second amplifier stage can be operated at a potential more positive than the plate of the preceding stage by the amount of the grid bias on the second amplifier stage, this direct connection between the plate of one stage and the grid of the succeeding stage can be used. Figure 10H illustrates an application of this principle in the coupling of a pentode amplifier stage to the grid of a "hot- cathode" phase inverter. In this arrangement the values of cathode, screen, and plate resistor in the pentode stage are chosen such that the plate of the pentode is at approximately 0. 3 times the plate supply potential. The succeeding phase- inverter stage then operates with conventional values of cathode and plate resistor (same value of resistance) in its normal manner. This type of phase inverter is described in more detail in the section to follow. Direct Coupling 6 -8 Phase Inverters It is necessary in order to excite the grids of a push -pull stage that voltages equal in amplitude and opposite in polarity be applied to the two grids. These voltages may be obtained through the use of a push -pull input transformer such as is shown in figure 10C. It is possible also, without the attendant bulk and expense of a push -pull input transformer, to obtain voltages of the proper polarity and www.americanradiohistory.com 116 Vacuum Tube Amplifiers phase through the use of a so- called phase inverter stage. There are a large number of phase inversion circuits which have been developed and applied Elut the three shown in figure 12 have been found over a period of time to be the most satisfactory from the point of view of the number of components required and from the standpoint of the accuracy with which the two out -of -phase voltages are held to the same amplitude with changes in supply voltage and changes in tubes. All of these vacuum tube phase inverters are based upon the fact that a 180° phase shift occurs within a vacuum tube between the grid input voltage and the plate output voltage. In certain circuits, the fact that the grid input voltage and the voltage appearing across the cathode bias resistor are in phase is used for phase inversion purposes. "Hot- Cathode" Figure 12A illustrates the hot Phase Inverter cathode type of phase inverter. This type of phase inverter is the simplest of the three types since it requires only one tube and a minimum of circuit components. It is particularly simple when directly coupled from the plate of a pentode amplifier stage as shown in figure 10H. The circuit does, however, possess the following two disadvantages: (1) the cathode of the tube must run at a potential of approximately 0.3 times the plate supply voltage above the heater when a grounded common heater winding is used for this tube as well as the other heater -cathode tubes in a receiver or amplifier: (2) the circuit actually has a loss in voltage from its input to either of the output grids-about 0.9 times the input voltage will be applied to each of these grids. This does represent a voltage gain of about 1.8 in total voltage output with respect to input (grid -to -grid output voltage) but it is still small with respect to the other two phase inverter circuits shown. Recommended component values for use with a 6J5 tube in this circuit are shown in figure 12A. If it is desired to use another tube in this circuit, appropriate values for the operation of that tube as a conventional amplifier can be obtained from manufacturer's tube data. The value of RL obtained should be divided by two, and this new value of resistance placed in the circuit as RL. The value of Rk from tube manual tables should then be used as Rkl in this circuit, and then the total of Rkl and Rk2 should be equal to RL. "Floating Paraphase" alternate type of phase inverter sometimes called the "floating paraphase" is illustrated in figure 12B. This circuit is quite often used with a 6N7 Phase Inverter An THE OA RADIO "HOT CATHODE, PHASE INVERTER ® "FLOATING PARAPHAS6'PHASE RL 47 INVERTER Cc ma RG CC.02 22011 11 22011 G= © CATHODE COUPLED PHASE INVERTER Figure 12 THREE POPULAR PHASE -INVERTER CIRCUITS WITH RECOMMENDED VALUES FOR CIRCUIT COMPONENTS tube, and appropriate values for the 6N7 tube in this application are shown. The circuit shown with the values given will give a voltage gain of approximately 21 from the input grid to each of the grids of the succeeding stage. It is capable of approximately 70 volts peak output to each grid. The circuit inherently has a small unbalance in output voltage. This unbalance can be eliminated, if it is required for some special application, by making the resistor Rgl a few per cent lower in resistance value than RB3. The circuit shown in figure 12C gives approximately one half the voltage gain from the input grid to either of the grids of the succeeding stage that would be obtained from a single tube of the same type operating as a conventional R -C amplifier stage. Thus, with a 6SN7 -GT tube as shown (two 6J5's in one Cathode -Coupled Phase Inverter www.americanradiohistory.com Vacuum HANDBOOK .01 R5 Tube Voltmeter 117 R6 I.R RP1 D.C. INPUT o Ec EP _i11Figure 14 DIRECT COUPLED D -C Figure 13 VOLTAGE DIVIDER PHASE INVERTER AMPLIFIER same amplitude as the output of V but of opposite phase. envelope) the voltage gain from the input grid to either of the output grids will be approximately 7-the gain is, of course, 14 from the input to both output grids. The phase characteristics are such that the circuit is commonly used in deriving push -pull deflection voltage for a cathode -ray tube from a signal ended input signal. The first half of the 6SN7 is used as an amplifier to increase the amplitude of the applied signal to the desired level. The second half of the 6SN7 is used as an inverter and amplifier to produce a signal of the same amplitude but of opposite polarity. Since the common cathode resistor, Rk, is not by- passed the voltage across it is the algebraic sum of the two plate currents and has the same shape and polarity as the voltage applied to the input grid of the first half of the 6SN7. When a signal, e, is applied to the input circuit, the effective grid- cathode voltage of the first section is Ae/2, when A is the gain of the first section. Since the grid of the second section of the 6SN7 is grounded, the effect of the signal voltage across Rk (equal to e/2 if Rk is the proper value) is the same as though a signal of the same amplitude but of opposite polarity were applied to the grid. The output of the second section is equal to Ae /2 if the plate load resistors are the same for both tube sections. commonly used phase inverter is shown in figure 13. The input section (V,) is connected as a conventional amplifier. The output voltage from V, is impressed on the voltage divider R, -R,. The values of R, and R, are in such a ratio that the voltage impressed upon the grid of V2 is 1/A times the output voltage of V where A is the amplification factor of V,. The output of Vt is then of the Voltage Divider Phase Inverter A D -C 6 -9 Amplifiers Direct current amplifiers are special types used where amplification of very slow variations in voltage, or of d -c voltages is desired. amplifier consists of a single A simple d -c tube with a grid resistor across the input terminals, and the load in the plate circuit. A simple d -c amplifier circuit is shown in figure 14, wherein the the grid of one tube is connected directly to the plate of the preceding tube in such a manner that voltage changes on the grid of the first tube will be amplified by the system. The voltage drop across the plate coupling resistor is impressed directly upon the grid of the second tube, which is provided with enough negative grid bias to balance out the excessive voltage drop across the coupling resistor. The grid of the second tube is thus maintained in a slightly negative position. The d -c amplifier will provide good low frequency response, with negligible phase distortion. high frequency response is limited by the shunting effect of the tube capacitances, as in the normal resistance coupled amplifier. A common fault with d -c amplifiers of all types is static instability. Small changes in the filament, plate, or grid voltages cannot be distinguished from the exciting voltage. Regulated power supplies and special balancing circuits have been devised to reduce the effects of supply variations on these amplifiers. A successful system is to apply the plate potential in phase to two tubes, and to apply the exciting signal to a push -pull grid Basic D -C Amplifier Circuit www.americanradiohistory.com 118 Vacuum Amplifiers Tube THE RADIO BALANCE CONTROL Figure 15 LOFTIN -WHITE D -C AMPLIFIER Figure 16 PUSH -PULL D -C AMPLIFIER WITH EITHER SINGLE -ENDED OR PUSH -PULL INPUT circuit configuration. If the two tubes are identical, any change in electrode voltage is balanced out. The use of negative feedback can also greatly reduce drift problems. The "Loftin -Whiter Circuit Two stages amplifier d -c may be arranged, so that their plate supplies are effectively in series, as illustrated in figure 15. This is known as a Loftin White amplifier. All plate and grid voltages may be obtained from one master power supply instead of separate grid and plate supplies. A push-pull version of this amplifier (figure 16) can be used to balance out the effects of slow variations in the supply voltage. 6 -10 Single -ended Triode Amplifiers Figure 17 illustrates five circuits for the operation of Class A triode amplifier stages. Since the cathode current of a triode Class Al (no grid current) amplifier stage is constant with and without excitation, it is common practice to operate the tube with cathode bias. Recommended operating conditions in regard to plate voltage, grid bias, and load impedance for conventional triode amplifier stages are given in the RCA Tube Manual, RC -16. It is possible, under certain conditions to operate singleended triode amplifier stages pentode and tetrode stages as well) with excitation of sufficient amplitude that current is taken by the tube on peaks. type of operation is called Class A2 and Extended Class A Operation (and grid grid This is characterized by increased plate -circuit efficiency over straight Class A amplification without grid current. The normal Class A1 amplifier power stage will operate with a plate circuit efficiency of from 20 per cent to perhaps 35 per cent. Through the use of Class A2 operation it is possible to increase this plate circuit efficiency to approximately 38 to 45 per cent. However, such operation requires careful choice of the value of plate load impedance, a grid bias supply with good regulation (since the tube draws grid current on peaks although the plate current does not change with signal), and a driver tube with moderate power capability to excite the grid of the Class A2 tube. Figures 17D and 17E illustrate two methods of connection for such stages. Tubes such as the 845, 849, and 304TL are suitable for such a stage. In each case the grid bias is approximately the same as would be used for a Class Al amplifier using the same tube, and as mentioned before, fixed bias must be used - along with an audio driver of good regulation preferably a triode stage with a 1:1 or step down driver transformer. In each case it will be found that the correct value of plate load impedance will be increased about 40 per cent over the value recommended by the tube manufacturer for Class A1 operation of the tube. Class A power amplifier operates in such a way as to amplify as faithfully as possible the waveform applied to the grid of the tube. Large power output is of more importance than high voltage amplification, consequently gain characteristics may be sacrificed in power tube design to obtain more important power handling capabilities. Class A power tubes, such as the 45, 2A3 and 6ÁS7 are characterized by a low amplification factor, high plate dissipation and relatively high filament emission. The operating characteristics of a Class A Operation Characteristics of a Triode Power Amplifier www.americanradiohistory.com A Triode Amplifier Characteristics HANDBOOK E5 -(0. 119 68 x Ebb) ll There Ebb is the actual plate voltage of the Class A stage, and µ is the amplification factor of the tube. pA IMPEDANCE 3- COUPLING 4- 5- ® TRANSFORMER COUPLING 6- 7- © IMPEDANCE -TRANSFORMER COUPLING 8- Locate the E5 bias point on the IP vs. Ep graph where the E5 bias line crosses the plate voltage line, as shown in figure 18. Call this point P. Locate on the plate family of curves the value of zero -signal plate current, I corresponding to the operating point, P. Locate 2 x 1, (twice the value of 1p) on the plate current axis (Y- axis). This point corresponds to the value of maximum signal plate current, imu. Locate point x on the d -c bias curve at zero volts (Eg = 0), corresponding to the value of imax. Draw a straight line(x - y) through points x and P. This line is the load resistance line. Its slope corresponds to the value of the load resistance. Load Resistance, (in ohms) RL Zsz RL - emu . ¡max - Cain - tmin where e is in volts, i is in amperes, and RL is in ohms. -e1AS = 0 TRANSFORMER COUPLING At OPERATION - AUTO TRANSFORMER -DIAS © e+e TO CLASS C LOAD CLASS Az MODULATOR WITH AUTO-TRANSFORMER COUPLING Figure 17 Output coupling arrangements for single -ended Class A triode audio -frequency power amplifiers. triode amplifier employing an output transformer- coupled load may be calculated from the plate family of curves for the particular tube in question by employing the following steps: 1- The load resistance should be approximately twice the plate resistance of the tube for maximum undistorted power output. Remember this fact for a quick check calculations. Calculate the zero -signal bias voltage on 2- (Eg). Multiply the zero - signal plate current, Ii,, by the operating plate voltage, Ep. If the plate dissipation rating of the tube is exceeded, it is necessary to increase the bias (E5) on the tube so that the plate dissipation falls within the maximum rating of the tube. If this step is taken, operations 2 through 8 must be repeated with the new value of 9- Check: +e FOR E5. 10- For maximum power output, the peak a -c grid voltage on the tube should swing to 2E5 on the negative cycle, and to zero bias on the positive cycle. At the peak of the negative swing, the plate voltage reaches emu and the plate current drops to iain On the positive swing of the grid signal, the plate voltage drops to envia and the plate current reaches ima. The power output of the tube is: Power Output (watts) Po - (imax - ¡min) x (emaz - emin) 8 where i is in amperes and e is in volts. 11- The second harmonic distortion generated in a single -ended Class A triode amplifier, expressed as a percentage of the fundamental output signal is: www.americanradiohistory.com 120 250 MN111 to W200 SOW. cc 7 . W ai11 ,50 a RADIO ou f : THE Amplifiers Tube 11 ä _ a1 11 I71 gIf/ I\ Vacuum Figure 19 Normal single -ended pentode or beam tetrad. audio- frequency power output stage. xi I.1 IMIN - 0 v - I 200 EMIN. 300 EG PLATE VOLTS 400 1 EMAX. AVERAGE PLATE CHARACTERISTICS p. =4.2 Rp / / 1 100 - 2A3 OHMS PLATE DISSIPATION =15 WATTS = BOO LOAD RESISTANCE RL - EMAZ I - EMIN. MAX.' IMIN. OHMS POWER OUTPUT Po (IMAX. -IMIN) IEMAX- Ei,) WATTS 8 - IP X 'MAX. Figure 100 PERCENT 18 Formulas for determining the operating conditions for a Class A triode single -ended audio frequency power output stage. A typical load line has been drawn on the average plate characteristics of a type 2A3 tube to illustrate the procedure. % 0. 9 Ep 2d harmonic = (imax - imin) IP iman and the power output is somewhat less than Ip 2 (x Ep x Ip 100) -imin 2 Figure 18 illustrates the above steps as applied to a single Class A 2A3 amplifier stage. 6 -11 The operating characteristics of pentode power amplifiers may be obtained from the plate family of curves, much as in the manner applied to triode tubes. A typical family of pentode plate curves is shown in figure 20. It can be seen from these curves that the plate current of the tube is relatively independent of the applied plate voltage, but is sensitive to screen voltage. In general, the correct pentode load resistance is about Operating Characteristics of a Pentode Power Amplifier SECOND HARMONIC DISTORTION IMAX.+ IMiN.) 2 plifier stage. Tubes of this type have largely replaced triodes in the output stage of receivers and amplifiers due to the higher plate efficiency (30 % -40 %) with which they operate. Tetrode and pentode tubes do, however, introduce a considerably greater amount of harmonic distortion in their output circuit, particularly odd harmonics. In addition, their plate circuit impedance (which acts in an amplifier to damp loudspeaker overshoot and ringing, and acts in a driver stage to provide good regulation) is many times higher than that of an equivalent triode. The application of negative feedback acts both to reduce distortion and to reduce the effective plate circuit impedance of these tubes. Single -ended Pentode Amplifiers Figure 19 illustrates the conventional circuit for a single -ended tetrode or pentode am- These formulae may be used for a quick check on more precise calculations. To obtain the operating parameters for Class A pentode amplifiers, the following steps are taken: 1- The imax point is chosen so as to fall on the zero -bias curve, just above the "knee" of the curve (point A, figure 20) . 2- A preliminary operating point, P, is determined by the intersection of the plate voltage line, Fp, and the line of imaz /2. www.americanradiohistory.com Push -Pull 'HANDBOOK Amplifiers 121 is: Power Output (watts) 6- The power output (¡max . ,iAA Po0 . .OAD_iNE t4/. ,OA^I¡vE AP=PB + 1.41 eR % P(STATIC VALUE) eMA% is equal to: ¡max ¡max Where IP Figure 20 % POWER AMPLIFIER is the negative control grid voltage at the operating point P The grid voltage curve that this point falls upon should be one that is about a the value of ER required to cut the plate current to a very low value (Point B). Point B represents imin on the plate current axis (y-axis). The line ima /2 should be located half-way between ima and twin trial load line is constructed about point P and point A in such a way that the lengths A -P and P -B are approximately equal. hen the most satisfactory load line has 4been determined, the load resistance may calculated: 3- A emax - envia imax - imin X RL ER + 0. 7 F.R. distortion is: distortion - min - 1, 2 x100 (Ix - Iy) - ¡min is the static plate current of + 1.41 of the tube. GRAPHIC DETERMINATION OF OPERATING CHARACTERISTICS OF A PENTODE RL 2d harmonic = PLATE VOLTS "V" -4)2 Where is the plate current at the point on the load line where the grid voltage, eR, is equal to: ER - 0. 7 F.R; and where Iy is the plate current at the point where 7- The percentage harmonic e MIN (Ix 32 l CHOOSE SOrHAT - ¡min) 5- The operating bias (ER) is the bias at point P. 3d harmonic (Ix - Ty) + 1.41 (1x - ly) ¡max -imin -1.41 ¡max - tmin 6 -12 x100 Push -Pull Audio Amplifiers A number of advantages are obtained through the use of the push -pull connection of two or four tubes in an audio -frequency power amplifier. Two conventional circuits for the use of triode and tetrode tubes in the push -pull connection are shown in figure 21. The two main advantages of the push -pull circuit arrangement are: (1) the magnetizing effect of the plate currents of the output tubes is cancelled in the windings of the output transformer; (2) even harmonics of the input signal (second and fourth harmonics primarily) generated in the push -pull stage are cancelled when the tubes are balanced. The cancellation of even harmonics generated in the stage allows the tubes to be oper- PUSH -PULL TRIODE AND TETRODE FIGURE distortion 21 www.americanradiohistory.com 122 Ii e.. itdNii, I.a r11\I,11 Vacuum 300 Amplifiers Tube THE J111111rrI RADIO 1111NN1111NNN11NNlli' NN111111. Ilun .rv IIM i iiii:iii:' : 11N / I . . 1/ "um= iGGiiGiiGiiG rM\RJ111N. 1111N1C::NII::..Ci ' I% 11111111NII Ci7 Ci GGG táiGGiiGiï N..llrCiC 1111I r A 250 , 11I.1111AIi 11 , VALUE Of ZERO SIGNAL PLATE CUR N/ W/N M Y w1'CCl M. :íRG /N!!:í. t1111P.. 50 111/111/11 !!I!!1!!1!!IIlII!I!!a 111111111111111 111111111111111111,¡m1111111111N111N 11111111111111111i1111111111111111111111 1111111111111111i/1111111111.1111111111 Iiill iiiiiii%1/N1111111111111111111 1111111111111 IIIIIIN11/I 1111111111 IIIINIIII111111 i'Il:il:?'i11NN11111111111111111111111 ..\! PLATE VOLTS 11 11111 1 1111111111 Cil:i=l. 'MIN_ 00 .1! -. II74VU'A l.... 11 o. I.M II 11111 P;11161111N111111111111111111111 1l:í1111D41N1 111111111111111111111111 70 e0 -30 -<0 30 -20 -0 0 -60 300 (EP) GRID VOLTS (EG) Figure 22 DETERMINATION OF OPERATING PARAMETERS FOR PUSH-PULL CLASS A TRIODE TUBES -in ated Class AB other words the tubes may be operated with bias and input signals of such amplitude that the plate current of alternate tubes may be cut off during a portion of the input voltage cycle. If a tube were operated in such a manner in a single -ended amplifier the second harmonic amplitude generated would be prohibitively high. Push -pull Class AB operation allows a plate circuit efficiency of from 45 to 60 per cent to be obtained in an amplifier stage depending upon whether or not the exciting voltage is of such amplitude that grid current is drawn by the tubes. If grid current is taken on input voltage peaks the amplifier is said to be operating Class AB2 and the plate circuit efficiency can be as high as the upper value just mentioned. If grid current is not taken by the stage it is said to be operating Class AB1 and the plate circuit efficiency will be toward the lower end of the range just quoted. In all Class AB amplifiers the plate current will increase from 40 to 150 per cent over the no- signal value when full signal is applied. The operating characteristics of push pull Class A amplifiers may also be determined from the plate family of curves for Operating Characteristics of Push -Pull Class A Triode Power Amplifier a particular triode tube by the following steps: 1- Erect a vertical line from the plate voltage axis (x -axis) at 0.6 Ep (figure 22), which intersects the Eg = 0 curve. This point of intersection (P), interpolated to the plate current axis (y -axis) may be taken as imp. It is assumed for simplification that imaz occurs at the point of the zero -bias curve corresponding to 2- 0.6 Ep. The power output obtainable from the two tubes is: Power output (Po) - i x Ep 5 where PO is expressed in watts, imax in amperes, and Ep is the applied plate voltage. 3- a preliminary load line through point P to the Ep point located on the x -axis (the zero plate current line). This load line represents % of the actual plate to -plate load of the Class A tubes. Therefore: Draw RL (plate -to- plate) www.americanradiohistory.com Ep = 4 x 1.6 ED max 0.6 Ep ¡Max Class I-ANDBOOK where RL is expressed in ohms, Ep in volts, and imu in amperes. Figure 22 illustrates the above steps applied to a push -pull Class A amplifier using two 2A3 tubes. 4- The average plate current is 0.636 Imp, and, multiplied by the plate voltage, Ep, will give the average watts input to the plates of the two tubes. The power output should be subtracted from this value to obtain the total operating plate dissipation of the two tubes. If the plate dissipation is excessive, a slightly higher value of RL should be chosen to limit the plate dissipation. 5- The correct value of operating bias, and the static plate current for the push -pull tubes may be determined from the Eg vs. 1p curves, which are a derivation of the Ep vs. Ip curves for various values of Eg. 6- The Fg vs. Ip curve may be constructed in this manner: Values of grid bias are read from the intersection of each grid bias curve with the load line. These points are transferred to the Eg vs. Ip graph to produce a curved line, A -B. If the grid bias curves of the Ep vs. Ip graph were straight lines, the lines of the Eg vs. 1p graph would also be straight This is usually not the case. A tangent to this curve is therefore drawn, starting at point A', and intersecting the grid voltage abscissa (x- axis). This intersection (C) is the operating bias point for fixed bias operation. 7- This operating bias point may now be plotted on the original Eg vs. 1p family of curves (C'), and the zero-signal current produced by this bias is determined. This operating bias point (C') does not fall on the operating load line, as in the case of a single -ended amplifier. 8- Under conditions of maximum power output, the exciting signal voltage swings from zero-bias voltage to zero -bias voltage for each of the tubes on each half of the signal cycle. Second harmonic distortion is largely cancelled out. 6 -13 B Audio Frequency Power Amplifiers Class The Class B audio- frequency power amplifier (figure 23) operates at a higher plate circuit efficiency than any of the previously described types of audio power amplifiers. Full- signal plate- circuit efficiencies of 60 to B Bt Audio DRIVER Amplifiers - BIAS Bt 123 MOD (GROUND FOR ZERO WAS OPERATING CONDITION) Figure 23 CLASS B AUDIO FREQUENCY POWER AMPLIFIER 70 per cent are readily obtainable with the tube types at present available for this type of work. Since the plate circuit efficiency is higher, smaller tubes of lower plate dissipation may be used in a Class B power amplifier of a given power output than can be used in any other conventional type of audio amplifier. An additional factor in favor of the Class B audio amplifier is the fact that the power input to the stage is relatively low under nosignal conditions. It is for these reasons that this type of amplifier has largely superseded other types in the generation of audio -frequency levels from perhaps 100 watts on up to levels of approximately 150,000 watts as required for large short -wave broadcast stations. Disadvantages of Class B Amplifier There are attendant dis advantageous features to the Operation operation of a power amplifier of this type; but all these disadvantages can be overcome by proper design of the circuits associated with the power amplifier stage. These disadvantages are: (1) The Class B audio amplifier requires driving power in its grid circuit; this disadvantage can be overcome by the use of an oversize power stage preceding the Class B stage with a step -down transformer between the driver stage and the Class -B grids. Degenerative feedback is sometimes employed to reduce the plate impedance of the driver stage and thus to improve the voltage regulation under the varying load presented by the Class B grids. (2) The Class B stage requires a constant value of average grid bias to be supplied in spite of the fact that the grid current on the stage is zero over most of the cycle but rises to values as high as one -third of the peak plate current on the peak of the exciting voltage cycle. Special regulated bias supplies have been used for this application, or B batteries can be used. However, a number www.americanradiohistory.com 124 Vacuum Tube Amplifiers THE of tubes especially designed for Class B audio amplifiers have been developed which require zero average grid bias for their operation. The 811A, 838, 805, 809, HY -5514, and TZ -40 are examples of this type of tube. All these so- called "zero- bias" tubes have rated operating conditions up to moderate plate voltages wherein they can be operated without grid bias. As the plate voltage is increased to to their maximum ratings, however, a small amount of grid bias, such as could be obtained from several 4 1/2-volt C batteries, is required. (3), A Class B audio -frequency power amplifier or modulator requires a source of plate supply voltage having reasonably good regulation. This requirement led to the development of the swinging choke. The swinging choke is essentially a conventional filter choke in which the core air gap has been reduced. This reduction in the air gap allows the choke to have a much greater value of inductance with low current values such as are encountered with no signal or small signal being applied to the Class B stage. With a higher value of current such as would be taken by a Class B stage with full signal applied the inductance of the choke drops to a much lower value. With a swinging choke of this type, having adequate current rating, as the input inductor in the filter system for a rectifier power supply, the regulation will be improved to a point which is normally adequate for a power supply for a Class B amplifier or modulator stage. Calculation of Operating Conditions of Class B Power Amplifiers The following procedure can be used for the calculation of the operating conditions of Class B power amplifiers when they are to operate into a resistive load such as the type of load presented by a Class C power amplifier. This procedure will be found quite satisfactory for the application of vacuum tubes as Class B modulators when it is desired to operate the tubes under conditions which are not specified in the tube operating characteristics published by the tube manufacturer. The same procedure can be used with equal effectiveness for the calculation of the operating conditions of beam tetrodes as Class AB2 amplifiers or modulators when the resting plate current on the tubes (no signal condition) is less than 25 or 30 per cent of the maximum -signal plate current. 1- With the average plate characteristics of the tube as published by the manufacturer before you, select a point on the Ep = E& (diode bend) line at about twice the plate current you expect the tubes to kick to under modulation. If beam tetrode tubes are concerned, select RADIO a point at about the same amount of plate current mentioned above, just to the right of the region where the Ib line takes a sharp curve downward. This will be the first trial point, and the plate voltage at the point chosen should be not more than about 20 per cent of the d -c voltage applied to the tubes if good plate- circuit efficiency is desired. Note down the value of ipp. and cp.,¡, at this point. 3- Subtract the value of epm¡ from the d -c plate voltage on the tubes. 4- Substitute the values obtained in the following equations: 2- P0 = pmau(Ebb RL_4 (Ebb epmin) = Power output from 2 tubes emu.) i pma: = Plate -to -plate load for 2 tubes Full signal efficiency (Nu) 78.5 Cl_evm Ebb /I Effects of Speech All the above equations are Clipping true for sine -wave operating conditions of the tubes concerned. However, if a speech clipper is being used in the speech amplifier, or if it is desired to calculate the operating conditions on the basis of the fact that the ratio of peak power to average power in a speech wave is approximately 4 -to-1 as contrasted to the ratio of 2 -to-1 in a sine wave-in other words, when non- sinusoidal waves such as plain speech or speech that has passed through a clipper are concerned, we are no longer concerned with average power output of the modulator as far as its capability of modulating a Class -C amplifier is concerned; we are concerned with its peak -power- output capability. Under these conditions we call upon other, more general relationships. The first of these is: It requires a peak power output equal to the Class -C stage input to modulate that input fully. The second one is: The average power output required of the modulator is equal to the shape factor of the modulating wave multiplied by the input to the Class -C stage. The shape factor of unclipped speech is approximately 0. 25. The shape factor of a sine wave is 0. 5. The shape factor of a speech wave that www.americanradiohistory.com eoo ï ó 600 -111 ma. U1 U Ò 400 125 200 O.C. - o N(Ong_. H-201111. agarla... - +6a vaLTs Ecc p10 N' been drawn on the overage characteristics of o type 811 tube. Parameters B EF e 6.3 VOLTS .u. Figure 24 Typical Class 8 o-f amplifier load line. The load line has m d' n NA :Ma ,C W! ..s/tll Class HANDBOOK 11iáse -O=e=sr = s I 400 600 1200 _ 2400 2000 1800 PLATE VOLTS (Ebb) AVERAGE PLATE CHARACTERISTICS TYPE 811 AND 811 -A has been passed through a clipper -filter arrangement is somewhere between 0. 25 and 0. 9 depending upon the amount of clipping that has taken place. With 15 or 20 db of clipping the shape factor may be as high as the figure of 0.9 mentioned above. This means that the audio power output of the modulator will be 90% of the input to the Class -C stage. Thus with a kilowatt input we would be putting 900 watts of audio into the Class -C stage for 100 per cent modulation as contrasted to perhaps 250 watts for unclipped speech modulation of 100 per cettt. Figure 24 shows a set of plate characteristics for a type 811A tube with a load line for Class B operation. Figure 25 lists a sample calculation for determining the proper operating conditions for obtaining approximately 185 watts output from a pair of the tubes with 1000 volts d -c plate potential. Also shown in figure 25 is the method of determining the proper ratio for the modulation transformer to couple between the 811's or 811A's and the anticipated final amplifier which is to operate at 2000 plate volts and 175 ma. plate current. Sample Calculation for 811A Tubes Lion shown in figure 25. or by reference to the published characteristics on the tubes to be used. (2) Determine the load impedance which will be presented by the Class C amplifier stage to be modulated by dividing the operating plate voltage on that stage by the operating value of plate current in amperes. (3) Divide the Class C load impedance determined in (2) SAMPLE CALCULATION CONDITION: 2 TYPE 811 TUBES, Ebb, = 1000 INPUT TO FINAL STAGE, 350 W. PEAR POWER OUTPUT NEEDED. 350 IS% = 370 FINAL AMPLIFIER Ebb = 2000 V. FINAL AMPLIFIER Ib = .175 A. FINAL AMPLIFIER ZL = -22SISL = 11400 R .175 EXAMPLE CHOSE POINT ON 811 TO RIGHT OF IP MAX. IG MAX. PEAK PO Ebb' F /G. EP MIN. A. EG MAX. _ RL = 4 X NP = 78.5 (1 .410 _ = X 900 - ) 1 (.9) 76.5 = WO (AVERAGE WITH SINE WAVE) WIN = - 260 Ió.5 Ib (MAXIMUM WO PEAR = = 369 W. = X80 = = 70.5 "b POIPEAR)_I813W W. WITH SINE WAVE) 100 DRIVING POWER 80 8800 n. = :9000 24 ) +100 A. .100 .410 0 (1000 -10o) = CHARACTERISTICS JUST Ecc. (PO /NT X. =.410 _ W. = = 260 MA e W WZ PR - W. TRANSFORMER: The method illustrated in figure 25 can be used in general for the determination of the proper transformer ratio to couple between the modulator tube and the amplifier to be modulated. The procedure can be stated as follows: (1) Determine the proper plate -to-plate load impedance for the modulator tubes either by the use of the type of calculaModulation Transformer Calculation 114 ZP sew TURNS RATIO = - 1.29 LA ZP = 1 29 = 1.14 STEP UP Figure 25 Typical calculation of operating conditions for a Class B a -f power amplifier using a pair of type 811 or 811A tubes. Plate characteristics and load line shown in figure 24. www.americanradiohistory.com 126 Vacuum Tube Amplifiers above by the plate -to -plate load impedance for the modulator tubes determined in (1) above. The ratio determined in this way is the sec ondary-to- primary impedance ratio. (4) Take the square root of this ratio to determine the secondary-to- primary turns ratio. If the turns ratio is greater than one the use of a step -up transformer is required. If the turns ratio as determined in this way is less than one a stepdown transformer is called for. If the procedure shown in figure '25 has been used to calculate the operating conditions for the modulator tubes, the transformer ratio calculation can be checked in the following manner: Divide the plate voltage on the modulated amplifier by the total voltage swing on the modulator tubes: 2 (Ebb e, 0). This ratio should be quite close numerically to the transformer turns ratio as previously determined. The reason for this condition is that the ratio between the total primary voltage and the d-c plate supply voltage on the modulated stage is equal to the turns ratio of the transformer, since a peak secondary voltage equal to the plate voltage on the modulated stage is required to modulate this stage 100 per cent. - Use of Clipper Speech Amplifier with Tetrode Modulator Tubes clipper speech amplifier is used in conjunction with a Class current. As stated previously, a Class B audio amplifier requires the driving stage to supply well -regulated audio power to the grid circuit of the Class B stage. Since the performance of a Class B modulator may easily be impaired by an improperly designed driver stage, it is well to study the problems incurred in the design of the driver stage. The grid circuit of a Class B modulator may be compared to a variable resistance which decreases in value as the exciting grid voltage is increased. This variable resistance appears across the secondary terminals of the driver transformer so that the driver stage is Class B Modulators RADIO called upon to deliver power to a varying load. For best operation of the Class 13 stage, the grid excitation voltage should not drop as the power taken by the grid circuit increases. These opposing conditions call for a high order of voltage regulation in the driver stage plate circuit. In order to enhance the voltage regulation of this circuit, the driver tubes must have low plate resistance, the driver transformer must have as large a step -down ratio as possible, and the d-c resistance of both primary and secondary windings of the driver transformer should be low. The driver transformer should reflect into the plate circuit of the driver tubes a load of such value that the required driving power is just developed with full excitation applied to the driver grid circuit. If this is done, the driver transformer will have as high a stepdown ratio as is consistent with the maximum drive requirements of the Class B stage. If the step -down ratio of the driver transformer is too large, the driver plate load will be so high that the power required to drive the Class B stage to full output cannot be developed. If the step-down ratio is too small the regulation of the driver stage will be impaired. When a B modulator stage, the plate current on that stage will kick to a higher value with modulation(due to the greater average power output and input) but the plate dissipation on the tubes will ordinarily be less than with sine -wave modulation. However, when tetrode tubes are used as modulators, the screen dissipation will be much greater than with sine -wave modulation. Care must be taken to insure that the screen dissipation rating on the modulator tubes is not exceeded under full modulation conditions with a clipper speech amplifier. The screen dissipation is equal to screen voltage times screen Practical Aspects of THE Driver Stage Calculations The parameters for the driver stage may be calculated from the plate characteristic curve, a sample of which is shown in figure 24. The required positive grid voltage (eg -m,$) for the 811A tubes used in the sample calculation is found at point X, the intersection of the load line and the peak plate current as found on the y -axis. This is + 80 volts. If a vertical line is dropped from point X to intersect the dotted current curves, it will be found that the current for a single 811A at this value of voltage is 100 milliamperes (point Y). peak grid driving power is therefore 80 x 0.100 = 8 watts. The approximate average driving power is 4 watts. This is an approximate figure because the grid impedance is not constant over the entire audio cycle. A pair of 2A3 tubes will be used as drivers, operating Class A, with the maximum excitation to the drivers occuring just below the point of grid current flow in the 2A3 tubes. The driver plate voltage is 300 volts, and the grid bias is -62 volts. The peak power developed in the primary winding of the driver transformer is: grid grid grid The Peak Power (Pe) = 2R1 I PE& ,a `Rp + RIwhere it is the amplification factor of the driver tubes (4.2 for 2A3). Eg is the peak grid swing of the driver stage (62 volts). Rp is the (watts) www.americanradiohistory.com Cathode Follower Amplifier HANDBOOK plate resistance of one driver tube (800 ohms). RL is % the plate -to -plate load of the driver stage, and Pp is 8 watts. Solving the above equation for RL, we obtain a value of 14,500 ohms load, plate -toplate for the 2A3 driver tubes. The peak primary voltage is: epri = 2RL x g Ft, +RL 493 volts and the turns ratio of the driver transformer (primary to % secondary) is: epri -= 493 = eg(ma:) 80 6.15:1 Plate Circuit One of the commonest distortion in a Class Impedance causes of B modu- lator is incorrect load impedance in the plate circuit. The purpose of the Class B modulation transformer is to take the power developed by the modulator (which has a certain operating impedance) and transform it to the operating impedance imposed by the modulated amplifier stage. If the transformer in question has the same number of turns on the primary winding as it has on the secondary winding, the turns ratio is 1:1, and the impedance ratio is also 1:1. If a 10,000 ohm resistor is placed across the secondary terminals of the transformer, a reflected load of 10,000 ohms would appear across the primary terminals. If the resistor is changed to one of 2376 ohms, the reflected primary impedance would also be 2376 ohms. If the transformer has twice as many turns on the secondary as on the primary, the turns ratio is 2:1. The impedance ratio is the square of the turns ratio, or 4:1. If a 10,000 ohm resistor is now placed across the secondary winding, a reflected load of 2,500 ohms will appear across the primary winding. Matching It can be seen from the above paragraphs that the Class B modulator plate load is entirely dependent upon the load placed upon the secondary terminals of the Class B modulation transformer. If the secondary load is incorrect, certain changes will take place in the operation of the Class B modulator stage. When the modulator load impedance is too low, the efficiency of the Class B stage is reduced and the plate dissipation of the tubes is increased. Peak plate current of the modulator stage is increased, and saturation of the modulation transformer core may result. "Talk -back" of the modulation trans- Effects of Plate Circuit Mis -match 127 former may result if the plate load impedance of the modulator stage is too low. When the modulator load impedance is too high, the maximum power capability of the stage is reduced. An attempt to increase the output by increasing grid excitation to the stage will result in peak -clipping of the audio wave. In addition, high peak voltages may be built up in the plate circuit that may damage the modulation transformer. 6 -14 Cathode- Follower Power Amplifiers The cathode -follower is essentially a power output stage in which the exciting signal is applied between grid and ground. The plate is maintained at ground potential with respect to input and output signals, and the output signal is taken between cathode and ground. Figure 26 illustrates four types of cathode - follower power amplifiers in common usage and figure 27 shows the output impedance (Ro), and stage gain (A) of both triode and pentode(or tetrode) cathode- follower stages. It will be seen by inspection of the equations that the stage voltage gain is always less than one, that the output impedance of the stage is much less than the same stage operated as a conventional cathode -return amplifier. The output impedance for conventional tubes will be somewhere between 100 and 1000 ohms, depending primarily on the transconductance of the tube. This reduction in gain and output impedance for the cathode -follower comes about since the stage operates as though it has 100 per cent degenerative feedback applied between its output and input circuit. Even though the voltage gain of the stage is reduced to a value less than one by the action of the degenerative feedback, the power gain of the stage (if it is operating Class A) is not reduced. Although more voltage is required to excite a cathode follower amplifier than appears across the load circuit, since the cathode "follows" along with the grid, the relative grid-to- cathode voltage is essentially the same as in a conventional amplifier. Types of Cathode- Follower Amplifiers Although the cathode -follower gives no voltage gain, it is an effective power amplifier where it is desired to feed a low- impedance load, or where it is desired to feed a load of varying impedance with a signal having good regulation. This latter capability Use of Cathode- Follower Amplifiers www.americanradiohistory.com 128 Vacuum Tube Amplifiers THE TRIODE -U ucr J,1 +1 Re (CATHODE PENTODE: Ro(cAr.,00E A = RADIO A + GM RL L +Rp (Rn,+Rea) RK, Rao RL ) RL(.U+I Ri +Rn2+ RL' R 1+RL Gu G.. Rea Figure 27 Equivalent factors for pentode (or tetrad.) cathode- follower power amplifiers. plifier tube, the components Figure 26 CATHODE-FOLLOWER OUTPUT CIRCUITS FOR AUDIO OR VIDEO AMPLIFIERS makes the cathode follower particularly effective as a driver for the grids of a Class B modulator stage. The circuit of figure 26A is the type of amplifier, either single -ended or push -pull, which may be used as a driver for a Class B modulator or which may be used for other applications such as feeding a loudspeaker where unusually good damping of the speaker is desired. If the d -c resistance of the primary of the transformer T2 is approximately the correct value for the cathode bias resistor for the am- Rk and Ck need not be used. Figure 26B shows an arrangement which may be used to feed directly a value of load impedance which is equal to or higher than the cathode impedance of the amplifier tube. The value of Cc must be quite high, somewhat higher than would be used in a conventional circuit, if the frequency response of the circuit when operating into a low- impedance load is to be preserved. Figures 26C and 26D show cathode -follower circuits for use with tetrode or pentode tubes. Figure 26C is a circuit similar to that shown in 26A and essentially the same comments apply in regard to the components Rk and Ck and the primary resistance of the transformer T2. Notice also that the screen of the tube is maintained at the same signal potential as the cathode by means of coupling capacitor Cd. This capacitance should be large enough so that at the lowest frequency it is desired to pass through the stage its reactance will be low with respect to the dynamic screen -tocathode resistance in parallel with Rd T2 in this stage as well as in the circuit of figure 26A should have the proper turns (or impedance) ratio to give the desired step -down or step -up from the cathode circuit to the load. Figure 26D is an arrangement frequently used in video systems for feeding a coaxial cable of relatively low impedance from a vacuum -tube amplifier. A pentode or tetrode tube with a cathode imped*tce as a cathode follower (1 /G,a) approximately the same as the cable impedance should be chosen. The 6AG7 and 6AC7 have cathode impedances of the same order as the surge impedances of certain types of low- capacitance coaxial cable. An arrangement such as 26D is also usable for feeding coaxial cable with audio or r -f energy where it is desired to transmit the output signal over moderate distances. The resistor Rk is added to the circuit as shown if the cathode impedance of the tube used is lower than the www.americanradiohistory.com HANDBOOK Feedback characteristic impedance of the cable. If the output impedance of the stage is higher than the cable impedance a resistance of appropriate value is sometimes placed in parallel with the input end of the cable. The values of Cd and Rd should be chosen with the same considerations in mind as mentioned in the discussion of the circuit of figure 26C above. INPUT SIGNAL ES The may 6 -15 Feedback Amplifiers It is possible to modify the characteristics of an amplifier by feeding back a portion of the output to the input. All components, circuits and tubes included between the point where the feedback is taken off and the point where the feedback energy is inserted are said to be included within the feedback loop. An amplifier containing a feedback loop is said to be a feedback amplifier. One stage or any number of stages may be included within the feedback loop. However, the difficulty of obtaining proper operation of a feedback amplifier increases with the bandwidth of the amplifier, and with the number of stages and circuit elements included within the feedback loop. The gain and phase shift of any amplifier are functions of frequency. For any amplifier containing a feedback loop to be completely stable the gain of such an amplifier, as measured from the input back to the point where the feedback circuit connects to the input, must be less than one Gain and Phase -shift in Feedback Amplifiers uTPUT E A VOLTAGE AMPLIFICATION WITH FEEDBACK 1 cathode follower conveniently be used asa method of coupling r -f or i -f energy between two units separated a considerable distance. In such an application a coaxial cable should be used to carry the r -f or i -f energy. One such application would be for carrying the output of a v -f -o to a transmitter located a considerable distance from the operating position. Another application would be where it is desired to feed a single -sideband demodulator, an FM adaptor, or another accessory with intermediate frequency signal from a communications receiver. A tube such as a 6CB6 connected in a manner such as is shown in figure 26D would be adequate for the i -f amplifier coupler, while a 6L6 or a 6AG7 could be used in the output stage of a v -f -o as a cathode follower to feed the coaxial line which carries the v -f-o signal from the control unit to the transmitter proper. AMPLIFIER GAIN= A FEEDBACK OR B PATH A The Cathode -Follower in R -F Stages 129 Amplifiers FEEDBACK IN DECIBELS = GAIN IN ABSENCE = B 8 -A OF B FEEDBACK FRACTION OF OUTPUT VOLTAGE FED BACK = NEGATIVE FOR NEGATIVE FEEDBACK IS 20 LOG (1 -A8) - 20 L0G MID FRED- GAIN WITHOUT FEEDBACK MIDFREO. GAIN WITH FEEDBACK DISTORTION WITH FEEDBACK RD _ DISTORTION WITHOUT FEEDBACK (1 8) -A RN _ 1 -Aa (1+ -) WHERE RD =OUTPUT IMPEDANCE OF AMPLIFIER WITH FEEDBACK RNA OUTPUT IMPEDANCE RL = OF AMPLIFIER WITHOUT FEEDBACK LOAD IMPEDANCE INTO WHICH AMPLIFIER OPERATES Figure 28 FEEDBACK AMPLIFIER RELATIONSHIPS at the frequency where the feedback voltage is in phase with the input voltage of the amplifier. If the gain is equal to or more than one at the frequency where the feedback voltage is in phase with the input the amplifier will oscillate. This fact imposes a limitation upon the amount of feedback which may be employed in an amplifier which is to remain stable. If the reader is desirous of designing amplifiers in which a large amount of feedback is to be employed he is referred to a book on the subject by H. W. Bode. Feedback may be either negative positive, and the feedback voltage may be proportional either to output voltage or output current. The most commonly used type of feedback with a -f or video amplifiers is negative feedback proportional to output voltage. Figure 28 gives the general operating conditions for feedback amplifiers. Note that the reduction in distortion is proportional to the reduction in gain of the amplifier, also that the reduction in the output impedance of the amplifier is somewhat greater than the reduction in the gain by an amount which is a function of the ratio of the Types of Feedback or H. W. Bode, fier Dsign,' and Feedback AmpliVan Nostrand Co., 250 Fourth Ave., "Network Analysis D. New York 3, N. Y. www.americanradiohistory.com 130 Vacuum THE Amplifiers Tube RADIO Figure 29 illustrates a very simple and effective application of negative voltage feedback to an output pentode or tetrode amplifier stage. The reduction in hum and distortion may amount to 15 to 20 db. The reduction in the effective plate impedance of the stage will be by a factor of 20 to 100 dependent upon the operating conditions. The circuit is commonly used in commercial equipment with tubes such as the 6SJ7 for VI and the 6V6 or 6L6 for V2. O° FEEDBACK 20 LOG e a 20L04 GAIN OP BOTH sTNGES R2 * R. RZ(G..V2 Ra +RR I ¡l I = RO) (VOLTAGE CAIN [ Goo, ( 6 -16 OrV2)) R2 ) ::.!-:1) RN ; R(G..vz Ro) 111 WHERE. RNR° RD = R2 OUTPUT R, %RD R, +R2 Rz GN.z Ro RCrLECTt0 LOAD IMPEDANCE .NEDPNCE RN = - RN ON V2 (USUALLY ABOUT S00 R) PEED°ACN RESISTOR R2 iRZ+RN(GwV2RO))(.+ PLATE IMPEDANCE R ) Vacuum -Tube Voltmeters The vacuum -tube voltmeter may be considered to be a vacuum -tube detector in which the rectified d -c current is used as an indication of the magnitude of the applied alternating voltage. The vacuum tube voltmeter (v.t.v.m.) consumes little or no power and it may be calibrated at 60 cycles and used at audio or radio frequencies with little change in the calibration. or V2 si mple v.t.v.m. is shown in figure 30. The plate load may be a mechanical device, such as a relay or a meter, or the output voltage may be developed across a resistor and used for various control purposes. The tube is biased by Ec and a fixed value of plate current flows, causing a fixed voltage drop across the plate load resistor, Rp. When a positive d -c voltage is applied to the input terminals it cancels part of the negative grid bias, making the grid more positive with respect to the cathode. This grid voltage change permits a greater amount of plate current to flow, and develops a greater voltage drop across the plate load resistor. A negative input voltage would decrease the plate current and decrease the voltage drop across Rp, The varying voltage drop across Rp may be employed as a control voltage for relays or other devices. When it is desired to measure various voltages, a voltage Basic Figure 29 SHUNT FEEDBACK CIRCUIT FOR PENTODES OR TETRODES This circuit requires only the addition of one resistor, R2, to the normal circuit for such an application. The plate impedance and distortion Introduced by the output stage are materially reduced. output impedance of the amplifier without feedback to the load impedance. The reduction in noise and hum in those stages included within the feedback loop is proportional to the reduction in gain. However, due to the reduction in gain of the output section of the amplifier somewhat increased gain is required of the stages preceding the stages included within the feedback loop. Therefore the noise and hum output of the entire amplifier may or may not be reduced dependent upon the relative contributions of the first part and the latter part of the amplifier to hum and noise. If most of the noise and hum is coming from the stages included within the feedback loop the undesired signals will be reduced in the output from the complete amplifier. It is most frequently true in conventional amplifiers that the hum and distortion come from the latter stages, hence these will be reduced by feedback, but thermal agitation and microphonic noise come from the first stage and wilt not be reduced but may be increased by feedback unless the feedback loop includes the first stage of the amplifier. D -C Vacuum Tube Voltmeter A Figure 30 SIMPLE VACUUM TUBE VOLTMETER www.americanradiohistory.com Vacuum Tube Voltmeters HANDBOOK 131 ZERO- ADJUST Figure 31 D -C VACUUM TUBE VOLTMETER Figure 32 BRIDGE -TYPE VACUUM TUBE VOLTMETER range switch (figure 31) may precede the v.t. v.m. The voltage to be measured is applied to voltage divider, R1, R2, R3, by means of the "voltage range" switch. Resistor R4 is used to protect the meter from excessive input voltage to the v.t.v.m. In the plate circuit of the tube an additional battery and a variable resistor ( "zero adjustment ") are used to balance out the meter reading of the normal plate current of the tube. The zero adjustment potentiometer can be so adjusted that the meter M reads zero current with no input voltage to the v.t.v.m. When a d -c input voltage is applied to the circuit, current flows through the meter, and the meter reading is proportional to the applied d -c voltage. The Bridge -type V.T.V.M. Another important use of a d-c amplifier is to show the exact point of voltages. This is done by means of a bridge circuit with two d -c amplifiers serving as two legs of the bridge (figure 32). With no input signal, and with matched triodes, no current will be read on meter M, since the IR drops across Ri and R2 are identical. When a signal is applied to one tube, the IR drops in the plate circuits become unbalanced, and meter M indicates the unbalance. In the same way, two d -c voltages may be compared if they are applied to the two input circuits. When the voltages are equal, the bridge is balanced and no current flows through the meter. If one voltage changes, the bridge becomes unbalanced and indication of this will be noted by a reading of the meter. balance between two d -c For the purpose of analysis, the operation of a modern v.t.v.m. will be described. The lleatbkit V -7A isa fit instrument for such adescription, since it is able to measure positive or negative d -c potentials, a -c r -m -s values, peak -to-peak values, and resistance. The circuit of this unit is shown in figure 33. A sensitive 200 d -c A Modern VTVM microammeter is placed in the cathode circuit of a 12AU7 twin triode. The zero adjust control sets up a balance between the twosections of the triode such that with zero input voltage applied to the first grid, the voltage drop across each portion of the zero adjust control is the same. Under this condition of balance the meter will read zero. When a voltage is applied to the first grid, the balance in the cathode circuits is upset and the meter indicates the degree of unbalance. The relationship between the applied voltage on the first grid and the meter current is linear and therefore the meter can be calibrated with a linear scale. Since the tube is limited in the amount of current it can draw, the meter movement is elec- tronically protected. The maximum test voltage applied to the 12AU7 tube is about 3 volts. Higher applied voltages are reduced by a voltage divider which has a total resistance of about 10 megohms. An additional resistance of 1- megohm is located in the d -c test prod, thereby permitting measurements to be made in high impedance circuits with minimum disturbance. The rectifier portion of the v.t.v.m. is shown in figure 34. When a -c measurements are desired, a 6AL5 double diode is used as a full wave rectifier to provide a d -c voltage pro portionalto the applied a-c voltage. This d -c voltage is applied through the voltage divider string to the 12AU7 tube causing the meter to indicate in the manner previously described. The a -c voltage scales of the meter are calibrated in both RMS and peak-to -peak values. In the 1.5, 5, 15, 50, and 150 volt positions of the range switch, the full a -c voltage being measured is applied to the input of the 6AL5 full wave rectifier. On the 500 and 1500 volt positions of the range switch, a divider network reduces the applied voltage in order to limit the voltage input to the 6AL5 to a safe recommended level. www.americanradiohistory.com 132 THE HEATHKIT PEAKTO -PEAK MODEL V -7A Figure VTVM 33 www.americanradiohistory.com RADIO Vacuum Tube Voltmeters HANDBOOK 02 A -C 150v 133 SNIELDED PROBE CASE 6AL5 INPUT 0---I 22 OM5 MAX MEG ff VTVM INPUT JACK TO DC COAA,L LINE A 7 MEG OOSi =PROBE TIP Ce-70 S Figure 34 Figure FULL -WAVE RECTIFIER FOR V.T.V.M. 35 -F PROBE SUITABLE FOR USE IN IKC -100 MC R RANGE The a -c calibrate control (figure 33) is used to obtain the proper meter deflection for the applied a -c voltage. Vacuum tubes develop a contact potential between tube elements. Such contact potential developed in the diode would cause a slight voltage to be present at all times. This voltage is cancelled out by proper application of a bucking voltage. The amount of bucking voltage is controlled by the a -c balance control. This eliminates zero shift of the meter when switching from a -c to d -c readings. For resistance measurements, a 1.5 volt battery is connected through a string of multipliers and the external resistance to be measured, thus forming a voltage divider across the battery, and a resultant portion of the battery voltage is applied to the 12AU7 twin triode. The meter scale is calibrated in resistance (ohms) for this function. Test Probes Auxiliary test probes may be used with the v.t.v.m. to extend the operating range, or to measure radio frequencies with high accuracy. Shown in figure 35 is a radio frequency probe which provides linear response to over 100 megacycles. A crystal diode is used as a rectifier, and d-c isolation is provided by a .005 uufd capacitor. The components of the detector are mounted within a shield at the end of a length of coaxial line, which terminates in the d-c input jack of the v.t.v.m. The readings obtained are RM1S, and should be multiplied by 1.414 to convert to peak readings. www.americanradiohistory.com CHAPTER SEVEN High Fidelity Techniques The art and science of the reproduction of sound has steadily advanced, following the major audio developments of the last decade. Public acceptance of home music reproduction on a "high fidelity" basis probably dates from the summer of 1948 when the Columbia L -P microgroove recording techniques were introduced. The term high fidelity refers to the reproduction of sound in which the different distortions of the electronic system are held below limits which are audible to the majority of listeners. The actual determination, therefore, of the degree of fidelity of a music system is largely psychological as it is dependent upon the ear and temperament of the listener. By and large, a rough area of agreement exists as to what boundaries establish a "hi -fi" system. To enumerate these boundaries it is first necessary to examine sound itself. As shown in figure 1, the sound wave of the fork has frequency, period, and pitch. The frequency is a measure of the number of vibrations per second of the sound. A fork tuned to produce 261 vibrations per second is tuned to the musical note of middle -C. It is of interest to note that any object vibrating, moving, or alternating 261 times per second will produce a sound having the pitch of middle -C. The pitch of a sound is that property which is determined by the frequency of vibration of the source, and not by the source itself. Thus an electric dynamo producing 261 c.p.s. will have a hum -pitch of middle -C, as will a siren, a gasoline engine, or other object having the same period of oscillation. ))111I) 7 -1 The Nature of Sound ))' III Experiments with a simple tuning fork in the seventeenth century led to the discovery that sound consists of a series of condensations and rarefactions of the air brought about by movement of air molecules. The vibrations of the prongs of the fork are communicated to the surrounding air, which in turn transmits the agitation to the ear drums, with the result that we hear a sound. The vibrating fork produces a sound of extreme regularity, and this regularity is the essence of music, as opposed to noise which has no such regularity. TUNING FORK Figure 1 VIBRATION OF TUNING FORK PRODUCES A SERIES OF CONDENSATIONS AND RAREFACTIONS OF AIR MOLECULES. THE DISPLACEMENT OF AIR MOLECULES CHANGES CONTINUALLY WITH RESPECT TO TIME, CREATING A SINE WAVE OF MOTION OF THE DENSITY VARIATIONS. 134 www.americanradiohistory.com Nature of Sound FREQUENCY (CYCLES NOTE D C EQUAL- TEMPERED 261.0 E F PER SECOND) G A B -I W I H C' 293.7 329.6 349.2 392.0 440.0 493.9 523.21 SCALE I Figure 2 EQUAL- TEMPERED SCALE CONTAINS TWELVE INTERVALS, EACH OF WHICH IS 1.06 TIMES THE FREQUENCY OF THE NEXT LOWEST. THE HALFINCLUDE THE INTERVALS TONE ABOVE NOTES PLUS FIVE ADDITIONAL NOTES: 277.2, 311.1, 370, 415.3, 466.2 REPRESENTED BY THE BLACK KEYS OF THE PIANO. THE 135 C 7 T/ME -J a. 2 Figure 3 THE COMPLEX SOUND OF A MUSICAL INSTRUMENT IS A COMBINATION OF SIMPLE SINE -WAVE SOUNDS, CALLED HARMONICS. THE SOUND OF LOWEST FREQUENCY IS TERMED THE FUNDAMENTAL. THE COMPLEX VIBRATION OF A CLARINET REED PRODUCES A SOUND SUCH AS SHOWN ABOVE. The Musical The musical scale is composed of notes or sounds of various frequencies that bear a pleasing aural relationship to one another. Certain combinations of notes are harmonious to the ear if their frequencies can be expressed by the simple ratios of 1:2, 2:3, 3:4, and 4:5. Notes differing by a ratio of 1:2 are said to be separated by an octave. The frequency interval represented by an octave is divided into smaller intervals, forming the musical scales. Many types of scales have been proposed and used, but the scale of the piano has dominated western music for the last hundred or so years. Adapted by J. S. Bach, the equal- tempered scale ( figure 2) has twelve notes, each differing from the next by the ratio 1:1.06. The reference frequency, or American Standard Pitch is A, or 440.0 cycles. Scale Harmonics and Overtones The complex sounds pro duced by a violin or a wind instrument bear little resemblance to the simple sound wave of the tuning fork. A note of a clarinet, for example (when viewed on an oscilloscope) resembles figure 3. Vocal sounds are even more complex than this. In 1805 Joseph Fourier advanced his monumental theorem that made possible a mathematical analysis of all musical sounds by showing that even the most complex sounds are made up of fundamental vibrations plus harmonics, or overtones. The tonal qualities of any musical note may be expressed in terms of the amplitude and phase relationship between the overtones of the note. To produce overtones, the sound source must be vibrating in a complex manner, such as is shown in figure 3. The resulting vibration is a combination of simple vibrations, producing a rich tone having fundamental, the octave tone, and the higher overtones. Any sound - - no matter how complex can be analyzed into pure tones, and can be reproduced by a group of sources of pure tones. The number and degree of the various harmonics of a tone and their phase relationship determine the quality of the tone. For reproduction of the highest quality, these overtones must be faithfully reproduced. A musical note of 523 cycles may be rich in twentieth order overtones. To reproduce the original quality of the note, the audio system must be capable of passing overtone frequencies of the order of 11,000 cycles. Notes of higher fundamental frequency demand that the audio system be capable of good reproduction up to the maximum response limit of the human ear, in the region of 15,000 cycles. Reproduction Many factors enter into the problem of high quality audio reproduction. Most important of these factors influence the overall design of the music system. These are: Restricted frequency range. Nonlinear distortions. Limitations 1- 23-Transient distortion. 4- Nonlinear frequency response. -Phase distortion. 6- Noise, "wow ", and "flutter ". 5 A restricted frequency range of reproduction will tend to make the music sound "tinny" and unrealistic. The fundamental frequency range covered by the various musical instruments and the human voice lies between 15 cycles and 9,000 cycles. Overtones of the instruments and the voice extend the upper audible limit of the music range to 15,000 cycles or so. In order to fully reproduce the musical tones falling within this range of frequencies the music system must be capable of flawlessly reproducing all frequencies within the range without discrimination. www.americanradiohistory.com THE RADIO High Fidelity Techniques 136 BASIC LIMITS FOR HIGH FIDELITY AND "GOOD QUALITY+ REPRODUCTION TYPE OF DISTORTION RESTRICTED FREQUENCY RANGE LIMIT HIGH FIDELITY 20 -IS000 CPS INTER MODULATION DISTORTION AT FULL OUTPUT 4 % 2 % HARMONIC DISTORTION AT FULL OUTPUT WOW" HUM AND NOISE "GOODY REPRODUCTI I SO -10000 10 W. S 0.1% -70 DB BELOW FULL OUTPUT CPS % 1W. -50 OB BELOW FULL OUTPUT Figure 4 Nonlinear qualities such as harmonic and intermodulation (IM) distortion are extremely objectionable and are created when the output of the music system is not exactly proportional to the input signal. Nonlinearity of any part of the system produces spurious harmonic frequencies, which in turn lead to unwanted beats and resonances. The combination of harmonic frequencies and intermodulation products produce discordant tones which are disagreeable to the ears. The degree of intermodulation may be measured by applying two tones f1 and f: of known amplitude to the input of the amplifier under test. The relative amplitude of the difference tone (f2-f1) is considered a measure of the intermodulation distortion. Values of the order of 4% IM or less define a high fidelity music amplifier. Response of the music system to rapid transient changes is extremely important. Transient peaks cause overloading and shock- excitation of resonant circuits, leaving a "hang- over" effect that masks the clarity of the sound. A system having poor transient response will not sound natural to the ear, even though the distortion factors are acceptably low. Linear frequency response and good power handling capability over the complete audio range go hand in hand. The response should be smooth, with no humps or dips in the curve over the entire frequency range. This requirement is particularly important in the electromechanical components of the music system, such as the phonograph pickup and the loudspeaker. Phase distortion is the change of phase angle between the fundamental and harmonic frequencies of a complex tone. The output wave envelope therefore is different from the envelope of the input wave. In general, phase distortion is difficult to hear in sounds having complex waveforms and may be considered to be sufficiently low in value if the IM figure of the amplifier is acceptable. Noise and distortions introduced into the program material by the music system must be kept to a minimum as they are particularly noticeable. Record scratch, turntable "rumble", and "flutter" can mar an otherwise high quality system. Inexpensive phonograph motors do not run at constant speed, and the slight variations in speed impart a variation in pitch ( wow) to the music which can easily be heard. Vibration of the motor may be detected by the pickup arm, superimposing a low frequency rumble on the music. The various distortions that appear in a music system are summarized in figure 4, together with suggested limits within which the system may truly be termed "high fidelity." 7 -2 The Phonograph The modern phonograph record is a thin disc made of vinylite or shellac material. Disc rotation speeds of 78.3, 33 1/3, and 45 r.p.m. are in use, with the older 78.3 r.p.m. speed gradually being replaced by the lower speeds. A speed of 16 2/3 r.p.m. is used for special "talking book" recordings. A continuous groove is cut in the record by the stylus of the recording machine, spiralling inward towards the center of the record. Amplitude variations in this groove proportional to the sound being recorded constitute the means of placing the intelligence upon the surface of the record. The old 78.3 r.p.m. recordings were cut approximately 100 grooves per inch, while the newer "micro- groove" recordings are cut approximately 250 grooves per inch. Care must be taken to see that the amplitude excursions of one groove do not fall into the adjoining groove. The groove excursions may be controlled by the system of recording, and by equalization of the recording equipment. The early commercial phonograph records were cut with a mechanical- acoustic system that produced a constant velocity characteristic with the amplitude of cut increasing as the recorded frequency decreased ( figure 5A) . When the recording technique became advanced enough to reproduce low audio frequencies, it was necessary to reduce the amplitude of the lower frequencies to prevent overcutting the record. A crossover point near 500 cycles was chosen, Recording Techniques www.americanradiohistory.com HANDBOOK High Fidelity Amplifier CONSTANT AMPLITUDE, 137 CROSSOVER FREQUENCY CONSTANT VELOCITY SURFACE NO /SE SURFACE NO /SE L__ 4 f 4 SO 100 200 -1- 1 4 500 1000 FREQUENCY 2000 5000 50 100 I 500 FREQUENCY (cvs) 1000 \ 2000 5000 ' (CPS) CONSTANT AMPLITUDE BELOW CROSSOVER FREQUENCY. CONSTANT VELOCITY ABOVE CROSSOVER FREQUENCY CONSTANT VELOCITY RECORDING V-- t 200 CROSSOVER FREQUENCY SURFACE NO SURFACE NOISE 50 100 200 1000 500 2000 5000 SO 200 100 500 FREQUENCY FREQUENCY (,P-) CONSTANT AMPLITUDE BELOW CROSSOVER FREQUENCY, NIGH FREQUENCY PRE -EMPHASIS ABOVE CROSSOVER FREQUENCY 1000 /SE 2000 solo° (cps) RESPONSE OF RECORD OF SC PLAYED ON PROPERLY COMPENSATED EQUIPMENT Figure 5 MODERN PHONOGRAPH RECORD EMPLOYS CONSTANT AMPLITUDE CUT BELOW CROSSOVER POINT AND HIGH FREQUENCY PRE- EMPHASIS (BOOST) ABOVE CROSSOVER FREQUENCY. "RIAA" PLAYBACK 10 20 40 70100 300 500 CURVE 1000 3000 10000 30000 FREQUENCY. CPS The most popular types of pickup cartridges in use today are the high impedance crystal unit, and the low impedance variable reluctance cartridge. The crystal pickup consists of a Rochelle salt element which is warped by the action of the phonograph needle, producing an electrical impulse whose frequency and amplitude are proportional to the modulation of the record groove. One of the new "transducer" crystal cartridges is shown in figure 6. When working into a high impedance load, the output of a high quality crystal pickup is of the The Phonograph Pickup and a constant amplitude groove was cut below this frequency ( figure 5B). This system does not reproduce the higher audio notes, since the recording level rapidly drops into the surface noise level of the record as the cutting frequency is raised. The modern record employs pre-emphasis of the higher frequencies to boost them out of the noise level of the record (figure 5C) . When such a record is played back on properly compensated equipment, the audio level will remain well above the background noise level, as shown in figure 5D. www.americanradiohistory.com 138 THE RADIO High Fidelity Techniques Figure 8 Figure 6 NEW CRYSTAL "TRANSDUCER" CARTRIDGE PROVIDES HIGH FIDELITY OUTPUT AT RELATIVELY HIGH LEVEL order of one -half volt or so. Inexpensive crystal units used in 78 r.p.m. record changers and ac -dc phonographs may have as much as two or three volts peak output. The frequency response of a typical high quality crystal pickup is shown in figure 7. The variable reluctance pickup is shown in figure 8. The reluctance of the air gap in a magnetic circuit is changed by the movement of the phonograph needle, creating a variable voltage in a small coil coupled to the magnetic lines of force of the circuit. The output impedance of the reluctance cartridge is of the order of a few hundred ohms, and the output is approximately 10 millivolts. For optimum performance, an equalized preamplifier stage is usually employed with the reluctance pickup. The circuit of a suitable unit is shown in figure 9. Equalization is provided by R5, R,, and C., with a low frequency crossover at about 500 cycles. Total equalization is 15 db. High frequency response may be limited by reducing the value of R. to 5,000 15,000 ohms. The standard records has a tip the microgroove has a tip radius - pickup stylus for 78 r.p.m. radius of .0025 inch, whereas (33 1/3 and 45 r.p.m.) stylus of .001 inch. Many pickups, "RELUCTANCE" CARTRIDGE STANDARD PICK -UP FOR IS MUSIC SYSTEM. Low stylus pressure of four grams insures minimum record wear. Dual stylus is used having two needle tip diameters for long playing and 78 R.P.M. recordings. therefore, are designed to have interchangeable cartridges or needles to accomodate the different groove widths. 7 -3 The High Fidelity Amplifier A block diagram of a typical high fidelity system is shown in figure 10. A preamplifier is used to boost the output level of the phonograph pickup, and to permit adjustment of input selection, volume, record compensation, and tone control. The preamplifier may be mounted directly at the phonograph turntable position, permitting the larger power amplifier to be placed in an out of the way position. The power amplifier is designed to operate from an input signal of a volt or so derived from the preamplifier, and to build this signal to the desired power level with a minimum amount of distortion. Maximum power output levels of ten to twenty watts are common for home music systems. The power supply provides the smoothed, d -c voltages necessary for operation of the preamplifier and power amplifier, and also the 6SC7 SNORT a w OUTPUT LEADS TO RELUCTANCE CARTRIDGE +10 Rts +5Rz 33M Z 0 a -5 20 r e R3 50 too Re 200 500 FREQUENCY 1000 2000 5000 68K 10000 C4 C5 T- e Rs 33K e+ 100 v HIGH PHONOGRAPH QUALITY CRYSTAL CARTRIDGE. (ELECTROVOICE 56 -DS POWER POINT TRANSDUCER) RESPONSE ,33K (cps) Figure 7 FREQUENCY Ce .0 RT 66 = TC3 _ .01 -10 Lai ¢ 66K 4 OF Figure 9 PREAMPLIFIER SUITABLE FOR USE WITH LOW LEVEL RELUCTANCE CARTRIDGE. www.americanradiohistory.com i HANDBOOK !PHONOGRAPH High Fidelity Amplifier 139 +20 RI f 1 M SPEAKER ENCLOSURE POWER SUPPLY -20 20 100 SO 200 SOO FREQUENCY I. Figure 10 BLOCK DIAGRAM OF HIGH FIDELITY MUSIC SYSTEM. 1000 2000 N TREBLE 5000 10000 (CeS) Figure 12 FREQUENCY RESPONSE CURVES FOR THE BASS AND TREBLE BOOST AND ATTENUATION CIRCUITS OF FIGURE 11. TREBLE BOOST AND ATTENUATION BASS BOOST AND ATTENUATION 1 SPONSE LOUD SPEAKER I v -t I POWER PREAMPLIFIER+ AMPLIFIER S TREBLE BOOST BASS BOOST TWEETERI INPUT RI eóosr C, Rz 80ó$r OUTPUT OUTPUT ATTENUATE ArrENUATE C2 Rn EQUIVALENT CIRCUITS EQUIVALENT CIRCUITS ATTENUATE ATTENUATE BOOST IN. Ri OUT. OUT. C2 Rz SIMPLE Equalizer networks are employed in high fidelity equipment to 1)- tailor the response curve of the system to obtain the correct compensate overall frequency response, 2) for inherent faults in the program material, merely to satisfy the hearing preference 3 of the listener. The usual compensation networks are combinations of RC and RL networks that provide a gradual attenuation over a given frequency range. The basic RC networks suitable for equlizer service are shown in figure 11. Shunt capacitance is employed for high frequency attenuation, and series capacitance is used for low frequency attenuation. A combination of these simple a -c voltage dividers may be used to provide almost any response, as shown in figure 12. It is common practice to place equalizers between two vacuum tubes in the low level stages of the preamplifier, as shown in figure 13. Bass and treble boost and attenuation of the order Tone Compensaton -to C1 our hold its own in the race for true fidelity. Speaker efficiency runs from about 10% for cone units to nearly 40% for high frequency tweeters. The frequency response of any speaker is a function of the design and construction of the speaker enclosure or cabinet that mounts the reproducer. O Figure 11 CIRCUITS MAY BE R -C USED FOR BASS AND TREBLE BOOST OR ATTENUATION. filament voltages (usually a -c) for the heaters of the various amplifier tubes. The loudspeaker is a device which couples the electrical energy of the high fidelity system to the human ear and usually limits the overall fidelity of the complete system. Great advances in speaker design have been made in the past years, permitting the loudspeaker to )-or +1zAx7 .04 E. OUTPUT 220 INPUT Figure 13 AND TREBLE LEVEL CONTROLS, AS EMPLOYED IN THE HEATHKIT WA -P2 PREAMPLIFIER. BASS B+ BASS CONTROL www.americanradiohistory.com TREBLE CONTROL K 140 THE RADIO High Fidelity Techniques 2N190 2N190 IRK 2N 190 12V 1K VARIABLE RELUCTANCE PICKUP e 3 9K OUTPUT or; TREBLE BASS Figure 14 TRANSISTORIZED HIGH- FIDELITY PREAMPLIFIER FOR USE WITH RELUCTANCE PHONOGRAPH CARTRIDGE. f100- Loudness Compensaron PAIN L EVEL +90 +BO +70 +60 +50 +40 +30 LOWER LIMIT OF HEARING +20 +10 0 10 20 50 100 200 500 FREQUENCY 1d00 2000 5000 10000 (CPS) 200 Figure 15 THE "FLETCHER -MUNSON" CURVE ILLUSTRATING THE INTENSITY RESPONSE OF THE HUMAN EAR. of 15 db may be obtained from such a circuit. simple transistorized preamplifier using this type of equalizing network is shown in figure 14. A The minimum threshold of hearing and the maximum threshold of pain vary greatly with the frequency of the sound as shown in the Fletcher- Munson curves of figure 15. To maintain a reasonable constant tonal balance as the intensity of the sound is changed it is necessary to employ extra bass and treble boost as the program level is decreased. A simple variable loudness control is shown in figure 16 which may be substituted for the ordinary volume control used in most audio equipment. The Power The power amplifier stage of the music system must supply driving power for the loudspeaker. Commercially available loudspeakers are low impedance devices which present a Amplifier mVIO o w +5 z o a O1 0 SPEAKER -5 RESPONSE ß: R -10 w aw o") V 14 ' O -RESONANT FREQUENCY OF -SPEAKER - BAFFLE COMBINATION a a m_ iiI Illi 20 RI -R2 -R 31 THREE SECTION POTENTIOMETER, /RC THE FOLLOWING, TYPE, BUILT OF R1 - lAC P011 -135 R2 -/RC MOLT /SECT /ON M13 -137 R3 - /RC MOLT /SECT /ON M13-128 10 20 50 100 SPEAKER 200 500 FREQUENCY Figure 16 VARIABLE LOUDNESS CONTROL FOR USE IN LOW IMPEDANCE PLATE CIRCUITS. MAY BE PURCHASED AS IRC TYPE LC -1 LOUDNESS CONTROL. 1000 2000 5000 4 10000 (CPS) Figure 17 IMPEDANCE AND FREQUENCY RESPONSE OF "4 -OHM" 12 -INCH SPEAKER PROPERLY MOUNTED IN MATCHING BAFFLE. www.americanradiohistory.com Ir 20000 HANDBOOK High Fidelity Amplifier 141 FEEDBACK RESISTOR VJ' Figure 18 TYPICAL TRIODE AMPLIFIER WITH FEEDBACK LOOP. * ° MATCHED varying load of two to nearly one hundred ohms to the output stage ( figure 17) . It is necessary to employ a high quality output transformer to match the loudspeaker load to the relatively high impedance plate circuit of the power amplifier stage. In general, push pull amplifiers are employed for the output stage since they have even harmonic cancelling properties and permit better low frequency response of the output transformer since there is no d -c core saturation effect present. To further reduce the harmonic distortion and intermodulation inherent in the amplifier system a negative feedback loop is placed around one or more stages of the unit. Frequency response is thereby improved, and the output impedance of the amplifier is sharply reduced, providing a very low source impedance for the loudspeaker. PAIR RESISTORS Shown in figure 18 is a basic push -pull triode amplifier, using inverse feedback around the power output and driver stage. A simple triode inverter is used to provide 180- degree phase reversal to drive the grid circuit of the power amplifier stage. Maximum undistorted power output of this amplifier is about 8 watts. A modification of the basic triode amplifier is the popular Williamson circuit (figure 19) developed in England in 1947. This circuit rapidly became the "standard of comparison" in a few short years. Pentode power tubes are connected as triodes for the output stage, and negative feedback is taken from the secondary of the output transformer to the cathode of the input stage. Only the most linear portion of the tube characteristic curve is used. Although that portion has been extended by higher than normal plate supply voltage, it FEEDBACK RESISTOR SK-ISK 6SN7GT 65N7GT 807 0.25 INPUT OUTPUT 22K NJ 30KK{ = 1 10 20 4-400V. AT 140 MA Figure 19 U. S. VERSION OF BRITISH OUTPUT AT LESS THAN "WILLIAMSON" AMPLIFIER PROVIDES 10 WATTS POWER 2 °o INTERMODULATION DISTORTION. 6SN7 STAGE DIRECT COUPLING. www.americanradiohistory.com USES 142 THE RADIO High Fidelity Techniques 807/5881 TO FEEDBACK CIRCUIT 0 25 FROM 05N7GT PHASE OUTPUT INVERTER 0.25 807/5881 NOTE, P/N CONNECT IONS ARE POR 807 TUBES f00V Figure 20 "ULTRA- LINEAR" CONFIGURATION OF WILLIAMSON AMPLIFIER DOUBLES POWER OUTPUT, AND REDUCES IM LEVEL. SCREEN TAPS ON OUTPUT TRANSFORMER PERMIT "SEMI -TETRODE" OPERATION. is only a fraction of the curve normally used in amplifiers. Thus a comparatively low output power level is obtained with tubes capable of much more efficient operation under less stringent requirements. With 400 volts applied to the output stage, a power output of 10 watts may be obained wtih less than 2% inter modulation distortion. A recent variation of the Williamson circuit involves the use of a tapped output transformer. The screen grids of the push -pull amplifier stage are connected to the primary taps, allowing operating efficiency to approach that of the true pentode. Power output in excess of 25 watts at less than 2% intermodulation dis- Figure 21 "BABY HI -FI" AMPLIFIER IS DWARFED BY 12 -INCH SPEAKER ENCLOSURE This miniature music system is capable of excellent performance in the small home or apartment. Preamplifier, bass and treble controls, and volume control are all incorporated in the unit. Amplifier provides 4 watts output at 4 IM distortion. tortion may be obtained with this circuit (figure 20) . 7 -4 Amplifier Construction Wiring Assembly and layout of high fidelity audio amplifiers follows the general technique described for other forms of electronic equipment. Extra care, however, must be taken to insure that the hum level of the amplifier is extremely low. A good hi -fi system has excellent response in the 60 cycle region, and even a minute quantity of induced a -c voltage will be disagreeably audible in the loudspeaker. Spurious eddy currents produced in the chassis by the power transformer are usually responsible for input stage hum. To insure the lowest hum level, the power transformer should be of the "upright" type instead of the "half-shell" type which can couple minute voltages from the windings to a steel chassis. In addition, part of the windings of the half -shell type project below the chassis where they are exposed to the input wiring of the amplifier. The core of the power transformer should be placed at right angles to the core of a nearby audio transformer to reduce spurious coupling between the two units to a Techniques minimum. It is common practice in amplifier design to employ a ground bus return system for all audio tubes. All grounds are returned to a single heavy bus wire, which in turn is grounded at one point to the metal chassis. This ground point is usually at the input jack of the amplifier. When this system is used, a -c chassis currents are not coupled into the amplifying stages. This type of construction is illustrated in the amplifiers described later in this chapter. www.americanradiohistory.com HANDBOOK Amplifier Construction NO FOR RI=e.2 220 .05 PHONO INPUT J, K^ e n SPEAKER 6AQ5 12AU7 12AÚ7 143 470K .05 e Sp MMr W R110 1,T2 len SPKR (HIGH LEVEL) 100 K M 3 1.e 1.9K SISOK R 10011 OS M 70 K BASE 5 6AQ5 VOLUME CONTROL CONTROL 1,7 TREBLE CONTROL +203 10 450V V. 1211 I +210V 6X5 TI CH1 3K 2W 2 115V ti IxÇ,A =4,5 9 12AÚ7 T,- 260 -0-260 ? 124Ú7 .20C111 =3 TOC1C 3 4 NOTES ALL RESISTORS 0.3 WATT UNLESS OTHERWISE SPECIFIED 2. ALL CAPACITOR VALUES IN MF UNLESS OTHERWISE SPECIFIED 3. RESISTORS MARKED A ARE MATCHED PAIRS 1. = 4 6.405 RAO5 Figure 22 SCHEMATIC, "BABY HI -FI" AMPLIFIER CH,-1.5 henry at 200 ma. Chicago Standard 6.3 volts at 4.0 volts at 90 ma., amp., upright mounting. Chicago -Standard PC -8420. T,-10 K, CT. to 8, 16 ohms. Peerless (Altec) S -510F. Care should be taken to reduce the capacitance to the chassis of high impedance circuits, or the high frequency response of the unit will suffer. Shielded "bath -tub" type capacitors should not be used for interstage coupling capacitors. Tubular paper capacitors are satisfactory. These should be spaced well away from the chassis. It is a poor idea to employ the chassis as a common filament return, especially for low level audio stages. The filament center-tap of the power transformer should be grounded, and twisted filament wires run to each tube socket. High impedance audio components and wiring should be kept clear of the filament lines, which may even be shielded in the vicinity of the input stage. In some instances, the filament center tap may be taken from the arm of a low resistance, wirewound potentiometer placed across the filament pins of the input tube socket. The arm of this potentiometer is grounded, and the setting of the control is adjusted for minimum speaker hum. 7 -5 The "Baby Hi Fi" A definite need exists for a compact, high fidelity audio amplifier suitable for use in the small home or apartment. Listening tests have shown that an average power level of less than C,A- B- C- 30 -20 -10 C -2327. Aid. 350 volt. Mallory Fp -330.7 NOTE -Feedback loop returns to 8 ohm tap on T, when 8 ohm speaker is used. one watt in a high efficiency speaker will provide a comfortable listening level for a small room, and levels in excess of two or three watts are uncomfortably loud to the ear. The "Baby Hi -Fi" amplifier has been designed for use in the small home, and will provide excellent quality at a level high enough to rattle the windows. Designed around the new Electro -Voice miniature ceramic cartridge, the amplifier will provide over 4 watts power, measured at the secondary of the output transformer. At this level, the distortion figure is below 1 %, and the IM figure is 4 %. At normal listening levels, the IM is much lower, as shown in figure 24. The Amplifier The schematic of the ampli fier is shown in figure 22. Bass and treble boost controls are incorporated in the circuit, as is the volume control. A dual purpose 12AU7 double triode serves as a voltage amplifier with cathode degeneration. A simple voltage divider network is used in the grid circuit to prevent amplifier overloading when the ceramic cartridge is used. The required input signal for maximum output is of the order of 0.3 volts. The output level of the Electro -Voice cartridge is approximately twice this, as shown in figure 7. The use of the high -level cartridge eliminCircuit www.americanradiohistory.com 144 THE RADIO High Fidelity Techniques Figure 23 UNDER -CHASSIS VIEW OF "BABY HI -FI" Low level audio stages are at upper left, with components mounted between socket pins and potentiometer controls. 6X5 socket is at lower cen- ter of photo with filter choke CH, at right. Feedback resistor R, is at left of rectifier socket. ates the necessity of high gain amplifiers required when low level magnetic pickup heads are used. Problems of hum and distortion introduced by these extra stages are thereby eliminated, greatly simplifying the amplifier. The second section of the 12AU7 is used for bass and treble boost. Simple R -C networks are placed in the grid circuit permitting gain boost of over 12 db at the extremities of the response range of the amplifier. A second 12AU7 is employed as a direct coupled "hot-cathode" phase inverter, capacitively coupled to two 6AQ5 pentode connected output tubes. The feedback loop is run from the secondary of the output transformer to the cathode of the input section of the phase inverter. The power supply of the "Baby Hi -Fi" consists of a 6X5 -GT rectifier and a capacitor input filter. A second R -C filter section is used to smooth the d -c voltage applied to the 12AU7 tubes. A cathode-type rectifier is used in preference to the usual filament type to prevent voltage surges during the warm -up period of the other cathode -type tubes. Amplifier The complete amplifier is built upon a small "amplifier foundation" chassis and cover measuring 5 "x7 "x6" (Bud CA- 1754). Height of the amplifier including dust cover is 6 ". The power transformer (T,) and output transformer (Tl) are placed in the rear corners of the chassis, with the'6X5 -GT rectifier socket placed between them. The small filter choke (CH,) is mounted to the wall of the chassis and may be seen in the under -chassis photograph of figure 23. The four audio tubes are placed in a row across the front of the chassis. Viewed from the front, the 12AU7 tubes are to the left, and the 6AQ5 tubes are to the right. The three section filter capacitor (C,A, B, C) is a chassis mounting unit, and is placed between the rectifier tube and the four audio tubes. Since the chassis is painted, it is important that good grounding points be made at each tube socket. The paint is cleared away Construction s- t o a 3 EQUIVALENT SINE WAVE WATTS Figure 24 INTERMODULATION CURVE FOR "BABY HI -Fl" AS MEASURED ON HEATHKIT INTERMODULATION ANALYZER. www.americanradiohistory.com HANDBOOK "Baby Hi -Fi" 145 Figure 25 TYPICAL INTERMODULATION TEST OF AUDIO AMPLIFIER. Audio tones of two frequencies are applied to input of amplifier under testi and amplitude of "sum" or "difference" frequency is measured, providing relative inter -modulation figure. beneath the socket bolt heads, and lock nuts are used beneath the socket retaining nuts to insure a good ground connection. All ground leads of the first 12AU7 tube are returned to the socket, whereas all grounds for the rest of the circuit are returned to a ground lug of filter capacitor G. Since the input level to the amplifier is of the order of one-half volt, the problem of chassis ground currents and hum is not so prevalent, as is the case with a high gain input stage. Phonograph -type coaxial receptacles are mounted on the rear apron of the chassis, serving as the input and output connections. The four panel controls (bass boost, treble boost, volume, and a -c on) are spaced equidistant across the front of the chassis. Amplifier Wiring The filament wiring should be done first. The center -tap of the filament winding is grounded to a lug of the 6X5 -GT socket ring, and the 6.3 volt leads from the transformer are attached to pins 2 and 7 of the same socket. A twisted pair of wires run from the rectifier socket to the right -hand 6AQ5 socket (figure 23). The filament leads then proceed to the next 6AQ5 socket and then to the two 12AU7 sockets in turn. The 12AU7 preamplifier stage is wired next. A two terminal phenolic tie -point strip is mounted to the rear of the chassis, holding the 12K decoupling resistor and the positive lead of the 10 µfd., 450 -volt filter capacitor. All B -plus leads are run to this point. Most of the components of the bass and treble boost system may be mounted between the tube socket terminals and the terminals of the two potentiometers. The feedback resistor R, is mounted between the terminal of the coaxial output connector and a phenolic tie -point strip placed beneath an adjacent socket bolt. When the wiring has been completed and checked, the amplifier should be turned on, and the various voltages compared with the values given on the schematic. It is important that the polarity of the feedback loop is correct. The easiest way to reverse the feedback polarity is to cross -connect the two plate leads of the 6AQ5 tubes. If the feedback polarization is incorrect, the amplifier will oscillate at a supersonic frequency and the reproduced signal will sound fuzzy to the ear. The correct connection may be determined with the aid of an oscilloscope, as the oscillation will be easily found. The builder might experiment with different values of feedback resistor RI, especially if a speaker of different impedance is employed. Increasing the value of R1 will decrease the degree of feedback. For an 8 -ohm speaker, Rt should be decreased in value to maintain the same amount of feedback. This amplifier was used in conjunction with General Electric S-1201A 12 -inch speaker mounted in an Electro -Voice KD6 Aristocrat speaker enclosure which was constructed from a www.americanradiohistory.com High Fidelity Techniques THE RADIO kit. The reproduction was extremely smooth, with good balance of bass and treble. requirements and the absence of expensive power and audio transformers it is more economical than conventional amplifiers of similar performance. 146 a 7 -6 A Transformerless 25 Watt Music Amplifier The Amplifier The output stage of this unusual amplifier is the single ended, push -pull type as shown in figure 27. The quiescent current is equal in both tubes with no d.c. current flowing through the speaker load. The absence of an output transformer allows 40 db of feedback to be app'ied by connecting the voice coil of the speaker directly to the cathode of the 12ÁT7 phase- inverter driver. In addition to its distortion reducing characteristic, the application of feedback serves to reduce the hum voltage which might otherwise be present. As the gain within the feedback loop is essentially unity, an additional voltage amplifier is used (with separate feedback) to build the input voltage up to the voice coil level. The power supply is a double half-wave selenium rectifier circuit developing +140 volts and -140 volts with respect to ground. The supply uses large filter capacitors, and no Circuit Because the output transformer is usually the weakest link in both frequency response and power output of an audio amplifier, several methods have been used to drive loudspeakers directly from the output tubes. These have either used non -conventional high -impedance loudspeakers, have been very inefficient, or have had low power output capabilities. The amplifier described in this section drives a conventional 16 ohm loudspeaker with normal class A amplifier efficiency, and supplies 25 watts of low distortion output throughout the audio range (figure 26). The amplifier requires an input signal of approximately one volt to drive it to maximum output. The unit attains its high performance through the use of 40 db of inverse feedback. Because of the relatively simple power supply Figure 26. 25 -WATT TRANS FORMERLESS AMPLIFIER PROVIDES ULTIMATE IN LISTENING PLEASURE FOR THE "GOLDEN EAR." Amplifier employs three triode tubes in single-ended push-pull configuration for maximum fidelity. The output 6082 tubes are placed across right end of chassis. 65N7 phase inverter is at rear, center; and low level stages are at the front, center. To the left are the power supply filter capacitors. In this particular amplifier, the 40 ohm, 20 watt filadropping resistor and a iron core reactor was used in its place (left, rear corner of the chassis). Across front of chassis are (I. to r.): power switch, input lack, and output stage balancing potentiometer. ment was small www.americanradiohistory.com eliminated HANDBOOK Transformerless Amplifier 12 AT 7 L I a 147 2 A T7 40 ±T + 40 150 + 150 loo 150 47K K INPUT K 6082 6SN7 -GT 0.1 * 0.1 6082 ;ff-6 + Sl 6082 56 K 5 5M 56K 4- 40 + 150 00 100 2 6K* 5 8K 1611 OUTPUT 2 680 40 150 . IM 5 I 1.5M M M 1.8 10K K 1K 4 1M 39K 00 100 56K 10 K 4 4 4 1.2 M +250 VOLTS +,o VOLTS 9 4,5 65147 4062 6082 6062 12AT7 6 7 7 6 6 7 11Hf--` SR 1.5Kz 40 S6W 9 i 7 560 2.5 K 10 SI 115V1., NOTE: 1. O% 40/150 12 AT7 r n ONE SIDE OP 70 CHASSIS. LINE A. roo fOWW W o *140 óI SR2 +aoo GROUNDED of S RS SR 4 0.5 160 ï5S -140 VOLTS 10 A A1° % T % REACTANCE NETWORKS 2. RESISTORS MARRED R ARE MATCHED PA /RS. 4.SRI. Qx P300500 ALL RESISTORS I -WATT UNLESS OTHERWISE NOTED. 3. CAPACITOR VALUES VOLTS GIVEN IN //FD. 75 MA., I50 VOLT. 5.SRz, SR3 =500MA., I I 15 150 VOLT. 150 SR4 8.29 10K 27 K 12K Figure 27. SCHEMATIC OF 25 -WATT MUSIC AMPLIFIER extra filtering is required. The output impedance of the supply is extremely low. To obtain higher voltage for the low level stages, additional selenium rectifiers are used in a voltage- adding configuration to obtain +250 and -250 volts. about -70 volts, but for this class of service the bias is held at -60 volts. A bias control is provided for one set of tubes so that the d.c. current flowing in the tubes may be equalized, and to insure that no d.c. current flows through the speaker voice coil. Circuit Details The type 6082 tube is not rated for use with fixed bias unless a limiting resistor is added in either the plate or the cathode circuit. Although this circuit does not use such resistors, their omission is feasible only because the tubes are used under quiescent conditions well below maximum ratings. With tubes of this type, it may be expected that the average current through the voice coil will drift with time but the presence of this un- The complete schematic of this amplifier is given in figure 27. Three type 6082 double triodes are employed in the output stage. These are 26.5 volt versions of the popular 6AS7G. These tubes are capable of 700 milliamperes of peak plate current per triode section at the plate voltage employed. The choice of the 6082 is an economy measure to allow the use of a series heater string. These tubes cut off at www.americanradiohistory.com High Fidelity Techniques THE RADIO balance current will generally be of little concern. In any event, the circuit has been designed so that the output stage can be conveniently rebalanced. voltage than the upper group. In the first voltage amplifier, bias is obtained from unbypassed cathode resistors since the loss of gain can easily be tolerated. The phase inverter- driver, however, has fixed bias applied to the grid from the -140 volt supply, since maximum gain is desired within the main feedback loop. 148 The Voltage The low level stages are all operated Class A with conventional circuitry. A separate driver is needed for each side of the output circuit, as insufficient output is obtained from the phase inverter to drive the output tubes directly. One side of the phase inverter has a larger load than the other, since the input to the lower group of output tubes has the speaker impedance in the cathode. This causes degeneration and necessitates higher input Amplifier Figure 28. UNDER -CHASSIS VIEW OF TRANSFORMERLESS AMPLIFIER Output tube sockets are at left, with power supply components at right. Components of preamplifier stages are grouped about the center sockets, mounted between socket pins and phenolic tie -point strips. Line fuse is mounted on rear apron of chassis. The Power Supply The high current power supply uses 300 µµfd. filter capacitors and 5 ohm protective resistors. R -C decoupling is used to minimize hum in the low level audio stages. As with all "power- transformerless" equipment, care must be taken when connecting this amplifier to other pieces of equipment to ensure that the grounded side of the power line is connected to the chassis. This may be achieved by the use of a polarized line plug, or a small isolation transformer may be employed. The Equalizing Circuit As would be expected, 40 db of feedback can only be applied within a loop having a minimum of phase shift or circuit instability will result. Since the loudspeaker O www.americanradiohistory.com HANDBOOK Transformerless Amplifier o 149 .0 0.8 0 6 -10 0. 0.2 20 10 100 1000 10 KC FREQUENCY 100 KC 1000 KC Figure 29. The balance adjustment for zero d.c. current through the speaker voice coil can be made with a milliammeter in series with the coil, or by measuring the voltage across the coil with a sensitive voltmeter. Amplifier The amplifier is built upon an aluminum chassis measuring 8" x 10" x 2 ". Perforated end pieces and 1/4 -inch holes drilled around the 6082 tube sockets insure adequate ventilation. Layout of the major components is shown in figure 26, and placement of the under -chassis components is shown in figure 28. As no a.c. power transformer is used, ground currents are of small concern, and the ground bus wiring technique need not be emConstruction 20 25 POWER OUTPUT (wnrrs) A- Overall frequency response of amplifier B-Distortion versus power output of amplifier impedance becomes inductive above the audio range it causes an increase in phase shift and loop gain. To avoid instability an impedance can be shunted across the voice coil to prevent the output reactance from rising at the higher audio frequencies. Three networks that have been used successfully for this purpose are shown in figure 27. The 180 ohm res for merely limits the maximum impedance of the output system and thus preven , excessive feedback. The 0.5 pfd. capacitor places a low impedance across the inductive load which is effective at the higher audio frequencies. The series 16 ohm resistor and 0.01 pfd. capacitor places a resistance across the speaker at the higher frequencies and an open circuit at the lower frequencies. This serves to provide constant impedance and feedback over the frequency range of the amplifier. 15 10 o 0 ployed. In its place, a tinned copper wire is run between the various chassis ground points. Ground connections may now be made to the socket grounding lugs, or to terminal strip ground points. A.c. filament and power leads are twisted wherever possible, and are run around the outer edges of the chassis. Point -to -point wiring technique is used, with small capacitors and resistors mounted to socket pins or to phenolic tie -point strips placed near the sockets. The small silicon rectifiers are mounted to tie -point strips placed near the upright filter capacitors. Several of the filter capacitors do not have their negative terminal at ground potential. It is therefore necessary to mount the capacitor on a phenolic plate and to slip a fiber insulating jacket over the metal shell. Amplifier The frequency response of the amplifier is flat within one db from 10 cycles to over 100 kilocycles. Since R -C coupled circuits are used throughout, there is no serious limitation on frequency response, and the response is down only 4 db at 250,000 cycles. The inter stage coupling networks limit the low frequency response below 10 cycles. Harmonic distortion and intermodulation at full rated output are exceptionally low and virtually independent of frequency. The ability to deliver 25 watts at 20 cycles and below with negligible distortion is practically impossible in a transformer -type circuit of similar mid -frequency power rating. Square wave response of the amplifier as measured between 20 cycles and 50 kilocycles is extremely good. Performance www.americanradiohistory.com --IilIFte LO-russ T seCT,oM LOIS r4ss i/ 011411 ..,f 7T 5e0110M VALUES SCALE FREQUENCY HIGH -PASS LOW -PASS .00 j-----; M1oM-491 T seCT,ON seCT,OM L LOAD RESISTANCE C 25.0 90 1000 10000 1100 9000 200 80 20.0 70 5 .5 1300 6 6 r .7 1400 1500 .6 e 60 15.0 9 .9 10 1.0 1600 1700 W tu 50 1600 1900 e000 Z- --r 1000 6000 W -1 u 5000 n 2000 M 4.s W 40 10.0 V Z 2500 < 2.0 O 4000 r u 9.0 O 3000 Z 3.0 e.0 3000 LC 7.0 40 .0 Ñ 50 5.0 u 4000 60 e.0 1h. W 2 6.0 70 90 5.0 20 7.0 I e0 100 9.0 10.0 4.0 --i- - J = 5000_4r 3 ta 1000 -- EA Ñ 8.0 I -5.- 15.0 C 1000 00 20.0 8000 300 30.0 150 3.0 J9000 2.5 10 10000 For both Pi -type 1000 - -- Courler, rrcdic e.0,9 9ni, FILTER DESIGN 2000 n 1000 ing CO. CHART and T -type Sections connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right side scale for high -pass) with load on left -hand side of right -hand scale by means of a straight -edge. Then read the value of L from the point where the edge intersects the left side of the center scale. Readings are in henries for frequencies in cycles per second. To find L, To find C, connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right side scale for high pass) with the load on the right -hand side of the right -hand scale. Then read the value of C from the point where the straightedge cuts the right side of the center scale. Readings are in microfarads for frequencies in cycles per d. For frequencies in kilocycles, C is expressed in thousands of micromicrofarads, L is expressed in mlllihenries. For frequencies in megacycles, L is expressed in microhenries and C is expressed in micromlcrofarads. For each tenfold increase In the value of load resistance multiply L by 10 and divide C by For each ten fold decrease in frequency multiply L by 10 end multiply C by 10. 150 www.americanradiohistory.com IO. CHAPTER EIGHT Radio Frequency Vacuum Tube Amplifiers TUNED RF VACUUM TUBE AMPLIFIERS Tuned r -f voltage amplifiers are used in receivers for the amplification of the incoming r -f signal and for the amplification of intermediate frequency signals after the incoming frequency has been converted to the intermediate frequency by the mixer stage. Signal frequency stages are normally called tuned r -f amplifiers and intermediate -frequency stages are called i-f amplifiers. Both tuned r -f and i -f amplifiers are operated Class A and normally operate at signal levels from a fraction of a microvolt to amplitudes as high as 10 to 50 volts at the plate of the last i -f stage in a receiver. first tuned circuit due to its equivalent coupled resistance at resonance. The noise voltage generated due to antenna radiation resistance and to equivalent tuned circuit resistance is similar to that generated in a resistor due to thermal agitation and is expressed by the following equation: En' k = R = Grid Circuit Considerations 1f = Since the full amplification of a receiver follows the first tuned circuit, the operating conditions existing in that circuit and in its coupling to the antenna on one side and to the grid of the first amplifier stage on the other are of greatest importance in determining the signal -to -noise ratio of the receiver on weak signals. highest ratio of signal -to -noise be impressed on the grid of the first r -f amplifier tube. Attaining the optimum ratio is a complex problem since noise will be generated in the antenna due to its equivalent radiation resistance (this noise is in addition to any noise of atmospheric origin) and in the First Tuned Circuit It is obvious that the 4kTRAf Where: E° = r -m -s value of noise voltage over the interval .1f T = 8 -1 = Boltzman's constant = 1374 X 10-22 joule per °K. Absolute temperature °K. Resistive component of impedance across which thermal noise is developed. Frequency band across which voltage is measured. In the above equation \f is essentially the frequency band passed by the intermediate frequency amplifier of the receiver under consideration. This equation can be greatly simplified for the conditions normally encountered in communications work. If we assume the following conditions: T = 300° K or 27° C or 80.5° F, room temperature; 1f = 8000 cycles (the average pass band of a communications receiver or speech amplifier) the equation remicrovolts. Acduces to: Et.m.s. = 0.0115 cordingly, the thermal -agitation voltage appearing in the center of half -wave antenna (assuming effective temperature to be 300° K) having a radiation resistance of 73 ohms is 151 www.americanradiohistory.com 152 R -F Vacuum Tube Amplifiers approximately 0.096 microvolts. Also, the thermal agitation voltage appearing across a 500,000 -ohm grid resistor in the first stage of a speech amplifier is approximately 8 microvolts under the conditions cited above. Further, the voltage due to thermal agitation being impressed on the grid of the first r -f stage in a receiver by a first tuned circuit whose resonant resistance is 50,000 ohms is approximately 2.5 microvolts. Suffice to say, however, that the value of thermal agitation voltage appearing across the first tuned circuit when the antenna is properly coupled to this circuit will be very much less than this value. It is common practice to match the impedance of the antenna transmission line to the input impedance of the grid of the first r -f amplifier stage in a receiver. This is the condition of antenna coupling which gives maximum gain in the receiver. However, when u -h -f tubes such as acorns and miniatures are used at frequencies somewhat less than their maximum capabilities, a significant improvement in signal -to -noise ratio can be attained by increasing the coupling between the antenna and first tuned circuit to a value greater than that which gives greatest signal amplitude out of the receiver. In other words, in the 10, 6, and 2 meter bands it is possible to attain somewhat improved signal -to -noise ratio by increasing antenna coupling to the point where the gain of the receiver is slightly reduced. It is always possible, in addition, to obtain improved signal -to -noise ratio in a v -h -f receiver through the use of tubes which have improved input impedance characteristics at the frequency in question over conventional types. The limiting condition for sensitivity in any receiver is the thermal noise generated in the antenna and in the first tuned circuit. However, with proper coupling between the antenna and the grid of the tube, through the first tuned circuit, the Noise Factor noise contribution of the first tuned circuit can be made quite small. Unfortunately, though, the major noise contribution in a properly designed receiver is that of the first tube. The noise contribution due to electron flow and due to losses in the tube can be lumped into an equivalent value of resistance which, if placed in the grid circuit of a perfect tube having the same gain but no noise would give the same noise voltage output in the plate load. The equivalent noise resistance of tubes such as the 6SK7, 6SG7, etc., runs from 5000 to 10,000 ohms. Very high Gm tubes such as the 6AC7 and 6AK5 have equivalent noise resistances as low as 700 to 1500 ohms. The lower the value of equivalent noise resistance, the THE RADIO lower will be the noise output under a fixed set of conditions. The equivalent noise resistance of a tube must not be confused with the actual input loading resistance of a tube. For highest signal -to -noise ratio in an amplifier the input loading resistance should be as high as possible so that the amount of voltage that can be developed from grid to ground by the antenna energy will be as high as possible. The equivalent noise resistance should be as low as possible so that the noise generated by this resistance will be lower than that attributable to the antenna and first tuned circuit, and the losses in the first tuned circuit should be as low as possible. The absolute sensitivity of receivers has been designated in recent years in government and commercial work by an arbitrary dimensionless number known as "noise factor" or N. The noise factor is the ratio of noise output of a "perfect" receiver having a given amount of gain with a dummy antenna matched to its input, to the noise output of the receiver under measurement having the same amount of gain with the dummy antenna matched to its input. Although a perfect receiver is not a physically realizable thing, the noise factor of a receiver under measurement can be determined by calculation from the amount of additional noise (from a temperature -limited diode or other calibrated noise generator) required to increase the noise power output of a receiver by a predetermined amount. Tube Input As has been mentioned in a pre - vious paragraph, greatest gain in a receiver is obtained when the antenna is matched, through the r -f coupling transformer, to the input resistance of the r -f tube. However, the higher the ratio of tube input resistance to equivalent noise resistance of the tube the higher will be the signal -to -noise ratio of the stage -and of course, the better will be the noise factor of the overall receiver. The input resistance of a tube is very high at frequencies in the broadcast band and gradually decreases as the frequency increases. Tube input resistance on conventional tube types begins to become an important factor at frequencies of about 25 Mc. and above. At frequencies above about 100 Mc. the use of conventional tube types becomes impracticable since the input resistance of the tube has become so much lower than the equivalent noise resistance that it is impossible to attain reasonable signal -to -noise ratio on any but very strong signals. Hence, special v -h-f tube types such as the 6AK5, 6ÁG5, and 6CB6 must be used. The lowering of the effective input resistLoading www.americanradiohistory.com HANDBOOK R ance of a vacuum tube at higher frequencies is brought about by a number of factors. The first, and most obvious, is the fact that the dielectric loss in the internal insulators, and in the base and press of the tube increases with frequency. The second factor is due to the fact that a finite time is required for an electron to move from the space charge in the vicinity of the cathode, pass between the grid wires, and travel on to the plate. The fact that the electrostatic effect of the grid on the moving electron acts over an appreciable portion of a cycle at these high frequencies causes a current flow in the grid circuit which appears to the input circuit feeding the grid as a resistance. The decrease in input resistance of a tube due to electron transit time varies as the square of the frequency. The undesirable effects of transit time can be reduced in certain cases by the use of higher plate voltages. Transit time varies inversely as the square root of the applied plate voltage. Cathode lead inductance is an additional cause of reduced input resistance at high frequencies. This effect has been reduced in certain tubes such as the 6S117 and the 6AK5 by providing two cathode leads on the tube base. One cathode lead should be connected to the input circuit of the tube and the other lead should be connected to the by -pass capacitor for the plate return of the tube. The reader is referred to the Radiation Laboratory Series, Volume 23: "Microwave Receivers" (McGraw -Hill, publishers) for additional information on noise factor and input loading of vacuum tubes. Amplifiers -F 153 OA AMPLIFICATION AT RESONANCE (APPROX.) =GMWLQ OB AMPLIFICATION AT RESONANCE (APPROX ) =GWMQ © AMPLIFICATION AT RESONANCE(APPRO[kGMK U) K2t-1-s 1 QP WHERE S PRI. ANO SEC. RESONANT AT SAME FREQUENCY 2 K IS COEFFICIENT OF COUPLING 1. IF FRI. AND SEC. Q ARE APPROXIMATELY THE SAME. 8 -2 TOTAL BANDWIDTH 1.2 K CENTER FREQUENCY MAXIMUM AMPLITUDE OCCURS AT CRITICAL COUPLING Plate- Circuit Considerations WHEN - KQP Noise is generated in a vacuum tube by the fact that the current flow within the tube is not a smooth flow but rather is made up of the continuous arrival of particles (electrons) at a very high rate. This shot effect is a source of noise in the tube, but its effect is referred back to the grid circuit of the tube since it is included in the equivalent noise resistance discussed in the preceding paragraphs. For the purpose of this section, it will be considered that the function of the plate load circuit of a tuned vacuum -tube amplifier is to deliver energy to the next stage with the greatest Plate Circuit Coupling efficiency over the required band of frequencies. Figure 1 shows three methods of inter stage coupling for tuned r -f voltage amplifiers. In figure IA omega (w) is 2n times the resonant frequency of the circuit in the plate of Figure Gain 1 equations for pentode r -f amplifier stages operating into a tuned load the amplifier tube, and L and Q are the inductance and Q of the inductor L. In figure 1B the notation is the same and M is the mutual inductance between the primary coil and the secondary coil. In figure 1C the notation is again the same and k is the coefficient of coupling between the two tuned circuits. As the coefficient of coupling between the circuits is increased the bandwidth becomes greater but the response over the band becomes progressively more double -humped. The response over the band is the most flat when the Q's of primary and secondary are approximately the same and the value of each Q is equal to 1.75/k. www.americanradiohistory.com 154 R -F Variable -Mu Tubes Vacuum Tube Amplifiers It is common practice to control the gain of a succession of r -f or i -f amplifier stages by varying the average bias on their control grids. H)wever, as the bias is raised above the operating value on a conventional sharp- cutoff tube the tube becomes increasingly non -linear in operation as cutoff of plate current is approached. The effect of such non -linearity is to cause cross modulation between strong signals which appear on the grid of the tube. When a tube operating in such a manner is in one of the first stages of a receiver a number of signals are appearing on its grid simultaneously and cross modulation between them will take place. The result of this effect is to produce a large number of spurious signals in the output of the receiver -in most in R -F Stages THE RADIO cases these signals will carry the modulation of both the carriers which have been cross modulated to produce the spurious signal. The undesirable effect of cross modulation can be eliminated in most cases and greatly reduced in the balance through the use of a variable -mu tube in all stages which have a -v-c voltage or other large negative bias applied to their grids. The variable -mu tube has a characteristic which causes the cutoff of plate current to be gradual with an increase in grid bias, and the reduction in plate current is accompanied by a decrease in the effective amplification factor of the tube. Variable -mu tubes ordinarily have somewhat reduced Gm as compared to a sharp- cutoff tube of the same group. Hence the sharp- cutoff tube will perform best in stages to which a-v -c voltage is not applied. RADIO- FREQUENCY POWER AMPLIFIERS All modern transmitters in the medium -frequency range and an increasing percentage of those in the v -h -f and u -h-f ranges consist of a comparatively low -level source of radio-frequency energy which is multiplied in frequency and successively amplified to the desired power level. Microwave transmitters are still predominately of the self- excited oscillator type, but when it is possible to use r -f amplifiers in s -h -f transmitters the flexibility of their application will be increased. The following portion of this chapter will be devoted, however, to the method of operation and calculation of operating characteristics of r-f power amplifiers for operation in the range of approximately 3.5 to 500 Mc. 8 -3 Class C R -F Power Amplifiers The majority of r -f power amplifiers fall into the Class C category since such stages can be made to give the best plate circuit efficiency of any present type of vacuum-tube amplifier. Hence, the cost of tubes for such a stage and the cost of the power to supply that stage is least for any given power output. Nevertheless, the Class C amplifier gives less power gain than either a Class A or Class B amplifier under similar conditions since the grid of a Class C stage must be driven highly positive over the portion of the cycle of the exciting wave when the plate voltage on the amplifier is low, and must be at a large negative potential over a large portion of the cycle so that no plate current will flow except when plate voltage is very low. This, in fact, is the fundamental reason why the plate circuit efficiency of a Class C amplifier stage can be made high -plate current is cut off at all times except when the plate -to- cathode voltage drop across the tube is at its lowest value. Class C amplifiers almost invariably operate into a tuned tank circuit as a load, and as a result are used as amplifiers of a single frequency or of a comparatively narrow band of frequencies. 2 shows the relation ships between the various voltages and currents over one cycle of the exciting grid voltage for a Class C amplifier stage. The notation given in figure 2 and in the discussion to follow is the same as given at the first of Chapter Six under "Symbols for Vacuum -Tube Parameters." The various manufacturers of vacuum tubes publish booklets listing in adequate detail alternative Class C operating conditions for the tubes which they manufacture. In addition, operating condition sheets for any particular type of vacuum tube are available for the asking from the different vacuum -tube manufacturers. It is, nevertheless, often desirable to determine optimum operating conditions for a tube under a particular set of circumstances. To assist in such calculations the following paragraphs are devoted to a method of calculating Class C operating conditions which is moderately simple and yet sufficiently accurate for all practical purposes. Relationships in Class C Stage www.americanradiohistory.com Figure Class HANDBOOK R C -F Amplifiers 155 tional grid voltage -plate current operating curves, the calculation is considerably simplified if the alternative "constant- current curve" of the tube in question is used. This is true since the operating line of a Class C amplifier is a straight line on a set of constantcurrent curves. A set of constant -current curves on the 250TH tube with a sample load line drawn thereon is shown in figure 5. In calculating and predicting the operation PLATE VOLTAGE EPM EBB P eP O1._.L.L- --'-- -1-I --- r I 1 PEAK PLATE CURRENT _ I If--t- - - --I-- -I I19P-.1.-9P+11 I I I I I I 7I I I t I 11 VI 1 I r I I EGM e - I I II II I I I I -I I - V t1 III - I III 11 III 0i- d11i Ecc -i FUNDAMENTAL COMPONENT OF PLATE CURRENT 'I I IG MAS. 1 : I { -1-,,,* I I - I I I 1------ I- I11 I III 111 I I III GRID III VOLTAGE 11-- -I G '-Ieca-1-- Figure Instantaneous electrode voltages and currents for I 2 and a amplifier Calculation of Class C Amplifier Operating Characteristics GRID I FI-CURRENT tank Class circuit C r -f power Although Class C opcrating conditions can be determined with the aid of the more conven- of a vacuum tube as a Class C radio -frequency amplifier, the considerations which determine the operating conditions are plate efficiency, power output required, maximum allowable plate and grid dissipation, maximum allowable plate voltage and maximum allowable plate current. The values chosen for these factors will depend both upon the demands of a par- ticular application and upon the tube chosen. The plate and grid currents of a Class C amplifier tube are periodic pulses, the durations of which are always less than 180 degrees. For this reason the average grid current, average plate current, power output, driving power, etc., cannot be directly calculated but must be determined by a Fourier analysis from points selected at proper intervals along the line of operation as plotted upon the constant- current characteristics. This may be done either analytically or graphically. While the Fourier analysis has the advantage of accuracy, it also has the disadvantage of being tedious and involved. The approximate analysis which follows has proved to be sufficiently accurate for most applications. This type of analysis also has the advantage of giving the desired information at the first trial. The system is direct in giving the desired information since the important factors, power output, plate efficiency, and plate voltage are arbitrarily selected at the beginning. first step in the method to described is to determine the power which must be delivered by the Class C amplifier. In making this determination it is well to remember that ordinarily from 5 to 10 per cent of the power delivered by the amplifier tube or tubes will be lost in well- designed tank and coupling circuits at frequencies below 20 Mc. Above 20 Mc. the tank and circuit losses are ordinarily somewhat above 10 per cent. The plate power input necessary to produce the desired output is determined by the plate efficiency: Pin = Pout/Np. For most applications tt is desirable to operate at the highest practicable efficiency. High efficiency operation usually requires less expensive tubes and power supplies, and the Method of The Calculation be www.americanradiohistory.com 156 R -F Vacuum Tube \ Amplifiers THE i'\ E \\ agi E \EE EE \u =\ MENE 7.0 E .0 E \1 O 7.0 E EEI \ .0 40 30 RADIO tt e RATIO ,. iáu t 3.0 I -10 EEE IMEN -20 -10 .1 RATIO Figure 3 Relationship between the peak value of the fundamental component of the tube plate current, and average plate current; as compared to the ratio of the instantaneous peak value of tube plate current, and average plate current amount of artificial cooling required is frequently less than for low- efficiency operation. On the other hand, high- efficiency operation usually requires more driving power and involves the use of higher plate voltages and higher peak tube voltages. The better types of triodes will ordinarily operate at a plate efficiency of 75 to 85 per cent at the highest rated plate voltage, and at a plate efficiency of 65 to 75 per cent at intermediate values of plate voltage. The first determining factor in selecting a tube or tubes for a particular application is the amount of plate dissipation which will be required of the stage. The total plate dissipation rating for the tube or tubes to be used in the stage must be equal to or greater than that calculated from: Pp = Pin - Pout. After selecting a tube or tubes to meet the power output and plate dissipation requirements it becomes necessary to determine from the tube characteristics whether the tube selected is capable of the desired operation and, if so, to determine the driving power, grid bias, and grid dissipation. The complete procedure necessary to determine a set of Class C amplifier operating conditions is given in the following steps: 1. Select the plate voltage, power output, and efficiency. Figure 4 Relationship between the ratio of the peak value of the fundamental component of the grid excitation voltage, and the overage grid bias; as compared to the ratio between instantaneous peak grid current and average grid current 2. Determine plate input from: Pin = Pout/Np. 3 Determine plate dissipation Pp= Pin - Pout Pp must from: not exceed maximum rated plate dissipation for tube or tubes selected. 4. Determine average plate current from: lb = Pin /Ebb 5. Determine approximate ;p.a. from: 4.9 lb for Np = 0.85 tpmas 4.5 lb for Np = 0.80 tpmas = 4.0 'b for N = 0.75 tpmax= 3.51b for Np =0.70 tpmax 6. Locate = = the point on constant -current characteristics where the constant plate current line corresponding to the approximate ipmax determined in step 5 crosses the line of equal plate and grid voltages (diode line). Read epmin at this point. In a few cases the lines of constant plate current will inflect sharply upward before reaching the diode line. In these cases epmin should not be read at the diode line but at the point where the plate current line intersects a line drawn from the origin through these points of inflection. www.americanradiohistory.com FIRST TRIAL POINT -s, N Om 157 Constant Current Calculations HANDBOOK FINAL POINT EIMAC 250TH CONSTANT CURRENT CHARACTERISTICS .... sa: pP: . , r-;pze o ó -_-. 00 _ Ti ERE 7b. ....... ......... , MOO EGO= - 240 LOAD LINE x00 XOD Ebb =4-3500 PLATE VOLTAGE -VOLTS FIGURE 5 Active portion of the operating load line for an Eimoc 250TH Class C r -f power amplifier, showing first trial point and the final operating point 7. Calculate Epm from: Epm = Ebb - epmin 13. 8. Calculate the ratio Ipm /lb from: 1pm lb 2 Ecc 1 Epm Fos Calculate a new value for ipmax from the ratio found in step 9. tpm as = (ratio from step 9) lb Ecc f3p =2.32( Ipm Ib - - µ egmp) Ebb fi 1 X L for tetrodes, where camp cos O - En, J 1112 tt is the grid- screen amplification factor, and Ec2 is the d -c screen voltage. and 10. cos \ 1- cos 6p constant current characteristics for the values of epmin and ipmax determined in steps 6 Calculate the cosine of one -half the angle of plate current flow from: Epm Op X - cos Op for triodes. 11. Read egmp and igmax from the 12. 1 - Np Ebb 9. From the ratio of Ipm /Ib calculated in step 8 determine the ratio ipmax /Ib from figure 3. 10. Calculate the grid bias voltage from: 14. Calculate the peak fundamental grid excitation voltage from: Earn = egmp - Ecc 1.57) 15. Calculate the ratio Egm /Ecc for the val- www.americanradiohistory.com 158 R -F Vacuum Amplifiers Tube ues of Ecc and Egm found in steps 13 12. and 14. Bp=2.32(1.73-1.57)=0.37 ratio 13. 1 - - 0.37 X (_ 3240 [0.37 3500 240) 37 37 14. - 240 volts Egm = 240 - ( -240) = 480 volts grid 15. Egm /Ecc = 16. igmax /Ic = 5.75 (from figure 4) 17. le = 0.430/5.75 = 0.075 amp. (75 ma. grid current) 18. Pd = 0.9X480X0.075 = 32.5 watts driving power = tgmax Ratio from step Ecc 1 Calculate the average grid current from the ratio found in step 16, and the value of igmax found in step 11: IC cos RADIO (9p = 68.3°) 16. Read igmax /Ic from figure 4 for the Egm /Ecc found in step 15. 17. THE 16 swing 18. Calculate approximate grid driving pow- er from: Pd 0.9 Egmlc = Calculate grid dissipation from: Pa = Pa + Eccic 19. Pg must not exceed the maximum rated grid dissipation for the tube selected. Sample typical example of a Class C amplifier calculation is shown A in the example below. Reference 3, 4 and 5 in the calcula- is made to figures tion. 1. Desired power output -800 watts. 2. Desired plate voltage -3500 volts. Desired plate efficiency -80 per cent (Np = 0.80) Pin = 800 /0.80 = 1000 watts 3. The power output of any type of r -f ampli- fier is equal to: Ib - It is frequently of importance to know the value of load impedance into which a Class C amplifier operating under a certain set of conditions should operate. This is simply R L_ Epm /Ipm. In the case of the operating conditions just determined for a 250TH amplifier stage the value of load impedance is: RL _ Approximate ipmax = 0.285 X 4.5 = 1.28 ampere trial point) 7. Epm = 3500 volts (see figure - 260 = 3240 5 11. egmp = 240 volts igmax = 0.430 amperes 3240 Ipm .495 6600 ohms -Xlb lb of Amplifier Tank Circuit volts (Both above from final point on figure Epm Ipm = In order to obtain good plate tank circuit tuning and low radiation of harmonics from an amplifier it is necessary that the plate tank circuit have the correct Q. Charts giving compromise values of Q for Class C amplifiers are given in the chapter, Generation o/ R -F Energy. However, the amount of inductance Q 9. ipmax /Ib = 4.1 (from figure 3) ipmax = 0.285X4.1 = 1.17 -- -Ipm first 8. Ipm /lb = 2X0.80X3500/3240 = 5600/3240 = 1.73 10. - ratio determined in step 8 above (in this type of calculation) by multiplying this ratio times Pp = 1000 800 = 200 watts Use 250TH; max. Pp = 250w;µ = 37. 6. epmin = 260 2 IpmEpm /2 = Po Ipm can be determined, of course, from the 4. lb = 1000/3500 = 0.285 ampere (285 ma.) Max. Ib for 250TH is 350 ma. 5. - 240 = - (- 240X0.75) = 14.5 watts grid dissipation Max. Pg for 250TH is 40 watts 19. Pg = 32.5 Calculation 480/ 5) required for a specified tank circuit Q under specified operating conditions can be calculated from the following expression: www.americanradiohistory.com Class HANDBOOK R B Q Quick Method of Calculating Amplifier Plate Efficiency The plate circuit efficiency of a Class B or Class C r -f amplifier can be determined from the following facts. The plate circuit efficiency of such an amplifier is equal to the product of two factors, which is equal to the ratio of Epm to Ebb (F, = Epm /Ebb) and which is proportional to the one -half angle of plate current flow, 0p. A graph of F, against both 0, and cos Bp is given in figure 6. Either 0p or cos Bp may be used to determine F,. Cos 0p may be determined either from the procedure previously given for making Class C amplifier computations or it may be determined from the following expression: F F Bp = - Ecc + Ebb µEBm-Epm Example of Method is desired to know the one -half angle of plate current flow and It the plate circuit efficiency for an 812 tube operating under the following conditions which have been assumed from inspection of the data and curves given in the RCA Transmitting Tube Handbook HB -3: 1. Ebb = 1100 volts Ecc = -40 volts 1.' ....,.... .......,.... .......-.... ...\\... Ism = 2 ir X operating frequency = Tank inductance = Required tube load impedance = Effective tank circuit Q tank circuit Q of 12 to 20 is recommended for all normal conditions. However, if a balanced push -pull amplifier is employed the tank receives two impulses per cycle and the circuit Q may be lowered somewhat from the above values. cos DN,IIII.`....MM.... MN A 159 R\...... Ri, L RL Q Amplifiers -F aee1 F2 aee o. i.........,,.. I..........,. awl D ore ore ar211.........., .... ..... aM 0 , 10 20 30 PP IN 40 50 e0 70 e0 90 100 110 120 ELECTRICAL DEGREES . .». <.. .... . . .:,. ... <.,. >... -..,» I cos I AP Figure 6 Relationship between Factor F_ and the half -angle of plate current flow in an amplifier with sine -wave input and output voltage, operating at a grid -bias voltage greater than cut -off 5. Np= F, X F, = 0.91 X 0.79 = 0.72 (72 per cent efficiency) F, could be called the plate -voltage-swing efficiency factor, and F2 can be called the operating -angle efficiency factor or the maximum possible efficiency of any stage running with that value of half -angle of plate current flow. Np is, of course, only the ratio between power output and power input. If it is desired to determine the power input, exciting power, and grid current of the stage, these can be obtained through the use of steps 7, 8, 9, and 10 of the previously given method for power inis put and output; and knowing that 0.095 ampere the grid circuit conditions can be determined through the use of steps 15, 16, 17, 18 and 19. = 29 Ea. = 120 volts Epm = 1000 volts 2. F, 3. cos 0p= = Epm/Ebb = 8 -4 0.91 - 29 X 40 + 1100 29 X 120 - 1000 60 - 0.025 2480 4. F2= 0.79 (by reference to figure 6) Class B Radio Frequency Power Amplifiers Radio frequency power amplifiers operating under Class B conditions of grid bias and excitation voltage are used in two general types of applications in transmitters. The first general application is as a buffer amplifier stage where it is desired to obtain a high value of power amplification in a particular stage. A particular tube type operated with a given plate voltage will be capable of somewhat greater output for a certain amount of excitation power when operated as a Class B ampli- www.americanradiohistory.com 160 R -F Tube Vacuum Amplifiers Ecz= 7-400 l RADIO THE V. Ec3=0v. Eci= II Ec=+Go __A ___ rd. 1 }s0 Eci=+4o i I \t Eci= +20 Eci= I 1 i I I iii / c2,Ec=+ioo J-- Icz Ec=+w l00 200 300 400 I Fri, _,--.-500 G00 700 -20 Eci=-4o G0o am 000 um, PLATE VOLTS .Hr, v. A..... ._.... ..,.,. ... Figure 7 AVERAGE PLATE CHARACTERISTICS OF 813 TUBE fier than when operated as fier. a Class C ampli- Calculation of Calculation of the operating conditions for this type of Characteristics Class B r -f amplifier can be carried out in a manner similar to that described in the previous paragraphs, except that the grid bias voltage is set on the tube before calculation at the value: Ecc = Ebb /IL. Since the grid bias is set at cutoff the one-half angle of plate current flow is 900; hence cos is fixed at 0.00. The plate circuit efficiency for a Class B r -f amplifier operated in this manner can be determined in the following manner: Operating NP =78.5 ( -1 Epm "Class Linear" B plication. Calculation of Operoting Parameters for a Class B Linear Amplifier Ebb // The calculated. Then, with the exciting voltage reduced to one -half for the no- modulation condition of the exciting wave, and with the same value of load resistance reflected on the tube, the plate input and plate efficiency will drop to approximately one-half the values at the 100 per cent positive modulation peak and the power output of the stage will drop to onefourth the peak- modulation value. On the negative modulation peak the input, efficiency, and output all drop to zero. In general, the proper plate voltage, bias voltage, load resistance and power output listed in the tube tables for Class B audio work will also apply to Class B linear r -f ap- The second type of Class B r-f amplifier is the so- called Class Il linear amplifier which is often used in transmitters for the amplification of a single - sideband signal or a conventional amplitude- modulated wave. Calculation of operating conditions may be carried out in a manner similar to that previously described with the following exceptions: The first trial operating point is chosen on the basis of the 100 per cent positive modulation peak of the modulated exciting wave. The plate circuit and grid peak voltages and currents can then be determined and the power input and output 7 illustrates characteristic curves for an 813 Figure the tube. Assume the plate supply to be 2000 volts, and the screen supply to be 400 volts. To determine the operating parameters of this tube as a Class B linear r -f amplifier, the following steps should be taken: 1. The grid bias is chosen so that the resting plate current will produce approximately 1/3 of the maximum plate dissipation of the tube. The maximum dissipation of the 813 is 125 watts, so the bias is set to allow one -third of this value, or 42 watts of resting dissipation. At a plate potential of 2000 volts, a www.americanradiohistory.com HANDBOOK 2. Linear Amplifier Parameters plate current of 21 milliamperes will produce this figure. Referring to figure 7, a grid bias of -45 volts is approximately correct. A practical Class 13 linear r -f amplifier runs at an efficiency of about 66% at full output, the efficiency dropping to about 33% with an unmodulated exciting signal. In the case of single- sideband suppressed carrier excitation, a no- excitation condition is substituted for the unmodulated excitation case, and the linear amplifier runs at the resting or quiescent input of 42 watts with no exciting signal. The peak allowable power input to the 813 is: Input Peak Power (Wp) _ (watts) Plate Dissipation X 100 (100 - BO ` GO Ec3=ov. Eci=+ioov. ci=rsov., 40 ¡ Ec-raov. Rib_ Eu=+sov xo Ecr+20 o - _ - . 200 100 Eg VS. E x P Ep - 400 _ x -= 0.189 ampere 2000 0.189 1580 = = 0.5 x .189 6000 ohms 9. If a loaded plate tank circuit Q of 12 is desired, the reactance of the plate tank capacitor at the r e s on an t frequency The plate current flow of the linear amplifier is 1800, and the plate current pulses have a peak of 3.14 times the maximum signal current: 3.14 resistance is: epmin 379 Wp Ep - 0.5ipmax The maximum signal plate current is: tpmax = should be: Reactance (ohms) = 0.595 ampere RL -- Referring to figure 7, a current of 0.605 ampere (Point A) will flow at a positive grid potential of 60 volts and a minimum plate potential of 420 volts. The grid is biased at -45 volts, so a peak r -f grid voltage of 60+45 volts = 105 volts is required. The grid driving power required for the B linear stage may be found by the aid of figure 8. It is one -quarter the product of the peak grid current times the peak grid voltage: Class 0.02 X 105 Pp = 10. 500 ohms For an operating frequency of 4.0 Mc., the effective resonant capacity is: 106 C= = 6.28 11. x 4.0 x 80 µµtd. 500 The inductance required to resonate at 4.0 Mc. with this value of capacity is: 500 L - 0.53 6000 = 12 Q 6. 300 Figure 8 CHARACTERISTICS OF 813 TUBE 8. The plate load 100 = 379 watts RL 5. E PLATE VOLTS 33 4. ECO +400 V. -% plate efficiency) 125 3. 161 watt 6.28 = x 4.0 19.9 microhenries 4 7. The single tone power output of the 813 stage is: Pp = 78.5 (Ep - epmin) x Ip Pp = 78.5 (2000 - 420) x .189 = 235 watts Grid Circuit The maximum positive grid potential is 60 volts, and the peak r -f grid voltage is 105 volts. Required driving power is 0.53 watt. The equivalent grid resistance of this stage is: Considerations www.americanradiohistory.com 1. 162 R -F Tube Vacuum Rig (e5)2 - 1052 - 2XPg 2X0.53 Amplifiers As in the case of the Class B audio amplifier the grid resistance of the linear amplifier varies from infinity to a low value when maximum grid current is drawn. To decrease the effect of this resistance excursion, a swamping resistor should be placed across the grid tank circuit. The value of the resistor should be dropped until a shortage of driving power begins to be noticed. For this example, a resistor of 3,000 ohms is used. The grid circuit load for no grid current is now 3,000 ohms instead of infinity, and drops to 2400 ohms when maximum grid current is drawn. 3. is chosen for the grid tank. The capacitive reactance required A circuit of Q 15 is: -= 2400 X = 160 ohms 15 4. At 4.0 Mc. the effective capacity is: 106 = C= 248 µµEd. 6.28x4X154 5. The inductive reactance required to reso- nate the grid circuit at 4.0 Mc. is: 160 L= = 6.4 microhenries 6.28 x 4.0 6. substituting the loaded grid resistance figure in the formula in the first paragraph, the grid driving power is now found to be approximately 2.3 watts. By Screen Circuit Considerations Special 8 -5 - 10,400 ohms 2. THE reference to the plate characteristic curve of the By 813 tube, it can be seen that at a minimum plate potential of 500 volts, and a maximum plate current of 0.6 ampere, the screen current will be approximately 30 milliamperes, dropping to one or two milliamperes in the quiescent state. It is necessary to use a well -regulated screen supply to hold the screen voltage at the correct potential over this range of current excursion. The use of an electronic regulated screen supply is recommended. R RADIO -F Power Amplifier Circuits The r-f power amplifier discussions of Sections 8 -4 and 8 -5 have been based on the assumption that a conventional grounded- cathode or cathode -return type of amplifier was in question. It is possible, however, as in the case of a -f and low-level r -f amplifiers to use circuits in which electrodes other than the cathode are returned to ground insofar as the signal potential is concerned. Both the plate-return or cathode -follower amplifier and the grid- return or grounded -grid amplifier are effective in certain circuit applications as tuned r -f power amplifiers. Disadvantages of Grounded -Cothode Amplifiers An undesirable aspect of the operation of cathode return r -f power amplifiers using triode tubes is that such amplifiers must be neutralized. Principles and methods of neutralizing r -f power amplifiers are discussed in the chapter Generation of R -F Energy. As the frequency of operation of an amplifier is increased the stage becomes more and more difficult to neutralize due to inductance in the grid and plate leads of the tubes and in the leads to the neutralizing capacitors. In other words the bandwidth of neutralization decreases as the frequency is increased. In addition the very presence of the neutralizing capacitors adds additional undesirable capacitive loading to the grid and plate tank circuits of the tube or tubes. To look at the problem in another way, an amplifier that may be perfectly neutralized at a frequency of 30 Mc. may be completely out of neutralization at a frequency of 120 Mc. Therefore, if there are circuits in both the grid and plate circuits which offer appreciable impedance at this high frequency it is quite possible that the stage may develop a "parasitic oscillation" in the vicinity of 120 Mc. This condition of restricted range neutralization of r -f power amplifiers can be greatly alleviated through the use of a cathode return or grounded -grid r -f stage. The grounded grid amplifier has the following advantages: Grounded -Grid R-F Amplifiers 1. The output capacitance of a stage is reduced to approximately one -half the value which would be obtained if the same tube or tubes were operated as a conventional neutralized amplifier. 2. The tendency toward parasitic oscillations in such a stage is greatly reduced since the shielding effect of the control grid be- www.americanradiohistory.com HANDBOOK Amplifier Grounded Grid tween the filament and the plate is effective over a broad range of frequencies. The feedback capacitance within the stage 3. is the plate -to- cathode capacitance which is ordinarily very much less than the gridto -plate capacitance. Hence neutralization is ordinarily not required. If neutralization is required the neutralizing capacitors are very small in value and are cross connected between plates and cathodes in a push -pull stage, or between the opposite end of a split plate tank and the cathode in a single -ended stage. The disadvantages of a grounded -grid amplifier are: 1. A large amount of excitation energy is required. However, only the normal amount of energy is lost in the grid circuit of the amplifier tube; all additional energy over this amount is delivered to the load circuit as useful output. 2. The cathode of a grounded -grid amplifier stage is "hot" to r.f. This means that the cathode must be fed through a suitable impedance from the filament supply, or the secondary of the filament transformer must be of the low- capacitance type and adequately insulated for the r -f voltage which will be present. 3. A grounded -grid r -f amplifier cannot be plate modulated 100 per cent unless the output of the exciting stage is modulated also. Approximately 70 per cent modulation of the exciter stage as the final stage is being modulated 100 per cent is recommended. However, the grounded -grid r -f amplifier is quite satisfactory as a Class B linear r -f amplifier for single sideband or conventional amplitude modulated waves or as an amplifier for a straight c -w or FM signal. Figure 9 shows a simplified representation of a grounded -grid triode r -f power amplifier stage. The relationships between input and out put power and the peak fundamental components of electrode voltages and currents are given below the drawing. The calculation of the complete operating conditions for a grounded-grid amplifier stage is somewhat more complex than that for a conventional amplifier because the input circuit of the tube is in series with the output circuit as far as the load is concerned. The primary result of this effect is, as stated before, that considerably more power is required from the driver stage. The normal power gain for a g -g stage is from 3 to 15 depending upon the grid circuit conditions chosen for the output stage. The higher the grid bias and grid swing required on the 163 c- {E4NEnr) IPr PONEN OUTPUT TO LOAD POWER DEL,ORSED n POWER PROM DO UER TO LOAD TOTAL PONE, DELIVERED BY ,SORKD DIVE, EON 5Y OUTPUT TUNE (IPI+ I4r) E MD NO .!S E4r Z. (PP,o..N E4r IPr i E4 =IPr o, POOR, EPr IP r Irr Err OUTPUT TUBE Irr Or E4r lc SUPPLY I4r O,OrE4NIt E4r I.N I lc Figure 9 GROUNDED -GRID CLASS B OR CLASS C AMPLIFIER The equations in the above figure give the relationships between the fundamental com- ponents of grid and plate potential and current, and the power input and power output of the stage. An expression for the approximate cathode impedance is given output stage, the higher will be the requirement from the driver. Calculation of Operating Conditions of Grounded Grid R -F Amplifiers It is most convenient to determine the op- erating conditions for a Class B or Class C grounded -grid r -f power amplifier in a two -step process. The first step is to determine the plate- circuit and grid- circuit operating conditions of the tube as though it were to operate as a conventional cathode- return amplifier stage. The second step is then to add in the additional conditions imposed upon the operating conditions by the fact that the stage is to operate as a grounded -grid amplifier. For the first step in the calculation the procedure given in Section 8 -3 is quite satisfactory and will be used in the example to follow. Suppose we take for our example the case of a type 304TL tube operating at 2700 plate volts at a kilowatt input. Following through the procedure previously given: 1. Desired power output -850 watts Desired Plate voltage -2700 volts Desired plate efficiency -85 per cent (Np = 0.85) www.americanradiohistory.com 164 2. 3. 4. -F R Tube Amplifiers Vacuum P1° = 850/0.85 = F, 1000 watts 850 = 150 watts = 1000 Type 304TL chosen; max. P, watts, /I= 12. = 300 Ib = 1000/2700 = 0.370 ampere (370 ma.) 5. Approximate ipma, = 4.9 X 0.370 = 1.81 ampere 6. epm;n= 140 volts (from 30411. con- stant- current curves) 7. Epm = 2700 - 140 = 2560 volts 8. 1./lb 9. 0.85 X 2700/2560 = 1.79 = 2 X igmax/Ib = 4.65 (from igmax = 4.65 X 0.370 = 1.72 amperes 11. egmp = 140 volts 12. Cos Op = = Op = Epm /Ebb = 2560/2700 = 0.95 of 59° (from figure 6) = 0.90 Np = F, X F2 = 0.95 X 0.90 = Approx. 0.85 (85 per cent plate efficiency) Now, to determine the operating conditions as a grounded -grid amplifier we must also know the peak value of the fundamental components of plate current. This is simply equal to (Ipm /Ib) lb, or: Ipm = 1.79 X 0.370 = 0.660 amperes (from 4 and 8 above) The total average power required of the driver (from figure 9) is equal to Egmlpm /2 (since the grid is grounded and the grid swing appears also as cathode swing) plus Pd which is 27.5 watts from 18 above. The total is: 0.480 amperes = 2.32 (1.79 -1.57) = 525 X 0.660 Total drive = - 172.5 watts 2 figure 3) 10. igma, RADIO F2 for Op - P, THE 0.51 59° plus 27.5 watts or 200 watts Therefore the total power output of the stage is equal to 850 watts (contributed by the 304TL) plus 172.5 watts (contributed by the driver) or 1022.5 watts. The cathode driving impedance of the 30411. (again referring to figure 7) is approximately: Zk = 525/(0.660 + 0.116) = approximately 675 ohms. 1 13. Ecc- - X 1- 0.51 [0.51 ( ` _ 2560 - 140) 12 - Plate- Return or Cathode- Follower 2700 12 J -385 volts 14. Egm = 140 -( -385) = 525 15. Egm /Ecc = 16. igmax /Ia = approx. 8.25 (extrapolated volts -1.36 from figure 4) 17. la = 0.480/8.25 grid current) 18. Pd = 0.9 19. Pg = Max. X 525 X 27.5 P, = 0.058 (58 ma. d -c 0.058 = 27.5 watts -( -385 X 0.058) = 5.2 watts for 304TL is 50 watts We can check the operating plate efficiency of the stage by the method described in Section 8 -4 as follows: Power Amplifier Circuit R -F diagram, elec- trodepotentials and currents, an d operating conditions for a cathode- follower r -f power amplifier are given in figure 10. This circuit can be used, in addition to the grounded -grid circuit just discussed, as an r -f amplifier with a triode tube and no additional neutralization circuit. However, the circuit will oscillate if the impedance from cathode to ground is allowed to become capacitive rather than inductive or resistive with respect to the operating frequency. The circuit is not recommended except for v -h -f or u -h -f work with coaxial lines as tuned circuits since the peak grid swing required on the r -f amplifier stage is approximately equal to the plate voltage on the amplifier tube if high- efficiency operation is desired. This means, of course, that the grid tank must be able to withstand slightly more peak voltage than the plate tank. Such a stage may not be plate modulated unless the driver stage is modulated the same percentage as the final amplifier. However, such a stage may be used as an amplifier or modulated waves (Class B linear) or as a c -w or FM amplifier. www.americanradiohistory.com HANDBOOK POWER OUTPUT TO LOAD G EpM ( POWER FROM DRIVER TO LOAD TOTAL POWER FROM DRIVER_ T. PM. IGM) EPM IPM 2 EPM IOM 2 (EPM +eGMP) IGM ECU IGM z 2 (EPM + eGMP) AppROA ASSUMING IGM y I e IC c APPROA ZG APPROA. I GM O S (Ecc eoup) lc ( EPM+ eGMP I I Figure 10 CATHODE -FOLLOWER R -F Control Grid Dissipation in Grounded -Grid Stages Tetrode tubes maybe operated as grounded 1.0 IC POWER ABSORBED Or OUTPUT TUBE GRID AND BIAS SUPPLY. - 165 circuit. If a conventional filament transformer is to be used the cathode tank coil may consist of two parallel heavy conductors (to carry the high filament current) by-passed at both the ground end and at the tube socket. The tuning capacitor is then placed between filament and ground.lt is possible in certain cases to use two r -f chokes of special design to feed the filament current to the tubes, with a conventional tank circuit between filament and ground. Coaxial lines also may be used to serve both as cathode tank and filament feed to the tubes for v -h -f and u -h -f work. - POWER DELIVERED BV OUTPUT TUBE - Amplifier -G POWER AMPLIFIER Showing the relationships between the tube potentials and currents and the input and output power of the stage. The approximate grid impedance also is given. grid (cathode driven) amplifiers by tying the grid and screen together and operating the tube as a high -u triode (figure 11). Combined grid and screen current, however, is a function of tube geometry and may reach destructive values under conditions of full excitation. Proper division of excitation between grid and screen should be as the ratio of the screen -to -grid amplification, which is approximately 5 for tubes such as the 4 -250A, 4 -400A, etc. The proper ratio of grid /screen excitation may be achieved by tapping the grid at some point on the filament choke, as shown. Grid dissipation is reduced, Lut the overall level of excitation is increased about 30% over the value required for simple grounded -grid operation. The design of such an amplifier stage is design of a grounded -grid amplifier stage as far as the first step is concerned. Then, for the second step the operating conditions given in figure 10 are applied to the data obtained in the first step. As an example, take the 304TL stage previously described. The total power required of the driver will be (from figure 10) approximately (2700X0.58);1.8) /2 or 141 watts. Of this 141 watts 27.5 watts (as before) will be lost as grid dissipation and bias loss and the balance of 113.5 watts will appear as output. The total output of the stage will then be approximately 963 watts. essentially the same as the 4 -200A, 4 -400A, fFt' ORIVe RFC- RFC rl Cathode Tank for G -G or C -F Power Amplifier The cathode tank circuit for either a grounded-grid or cathode -follower r -f power amplifier may be a conventional tank circuit if the filament transformer for the stage is of the low-capacitance high- voltage type. Conventional filament transformers, however, will not operate with the high values of r -f voltage present in such a FIGURE II TAPPED FILAMENT CHORE REDUCES EXCESSIVE GRID DISSIPATION IN G -G CIRCUIT. RFC--TWOP CAC., TAP 71- e.a II WINDINGS OP 191C WIRE, ES TURN! DIAM. TOTAL LENGTH IS SIR INCHES. GRID TURNS PROM GROUND CND OP ONE WINDING. I- 1 vOLTS AT AMPERE!. (VOLTAGE DROP ACROSS I.a VOLTS ) RFC Is www.americanradiohistory.com -F Vacuum Tube THE ,'Amplifiers 166 R 8 -6 Class ABI Radio Frequency Power Amplifiers » yll i iiiiiis vis w w1i /j o PORTIONOF EG-tP Class Aß1 r -f amplifiers operate under such conditions of bias and excitation that grid current does not flow over any portion of the input cycle. This is desirable, since distortion caused by grid current loading is absent, and also because the stage is capable of high power gain. Stage efficiency is about i 91JJiT1-!ßr - MOSTLINRWA,-P CURVE .Ì*n 15111111111114 MIN RADIO A, SIGNAL 111 IN plate current operating angle of 2100 is chosen, as compared to 62% for Class 58% when a operation. The level of static (quiescent) plate current for lowest distortion is quite critical for Class AB1 tetrode operation. This value is determined by the tube characteristics, and is not greatly affected by the circuit parameters or operating voltages. The maximum d -c plate potential is therefore limited by the static dissipation of the tube, since the resting plate current figure is fixed. The static plate current of a tetrode tube varies as the 3/2 power of the screen voltage. For example, raising the screen voltage from 300 to 500 volts will double the plate current. The optimum static plate current for minimum distortion is also doubled, since the shape of the Eg -Ip curve does not change. In actual practice, somewhat lower static plate current than optimum may be employed without raising the distortion appreciably, and values of static plate current of 0.6 to 0.8 of optimum may be safely used, depending upon the amount of nonlinearity that can be tolerated. As with the class B linear stage, the minimum plate voltage swing of the class AB1 amplifier must be kept above the d-c screen potential to prevent operation in the nonlinear portion of the characteristic curve. A low value of screen voltage allows greater r -f plate voltage swing, resulting in improvement in plate efficiency of the tube. A balance between plate dissipation, plate efficiency, and plate voltage swing must be achieved for best linearity of the amplifier. B The 5 -Curve The perfect linear amplifier delivers a signal that is a replica of the input signal. Inspection of the plate characteristic curve of a typical tube will disclose the tube linearity under class A operating conditions (figure 12). The curve is usually of exponential shape, and the signal distortion is held to a small value by operating the tube well below its maximum output, and centering operation over the most linear portion of the characteristic curve. Figure 12 CURVE Eg -Ip Amplifier most operation is confined to linear portion of characteristic curve. The relationship between exciting voltage in a Class ABI amplifier and the r -f plate circuit voltage is shown in figure 13. With a small value of static plate current the lower portion of the line is curved. Maximum undistorted output is limited by the point on the line (A) where the instantaneous plate voltage down to the screen voltage. This "hook" in the line is caused by current diverted from the plate to the grid and screen elements of the tube. The characteristic plot of the usual linear amplifier takes the shape of an S-curve. The lower portion of the curve is straightened out by using the proper value of static plate current, and the upper portion of the curve is avoided by limiting minimum plate voltage swing to a point substantially above the value of the screen voltage. The approximate operacing parameters may be Linear Amplifier obtained from the Constant Current curves (Eg -Ep) or the Eg -Ip curves of the tube in question. An operating load line is first approximated. One end of the load line is determined by the d -c operating voltage of the tube, and the required static plate current. As a starting point, let the product of the plate voltage and current equal the plate dissipation of the tube. Assuming we have a 4 -400A tetrode, this end of the load line will fall on point A (figure 14). Plate power dissipation is 360 watts (3000V @ 120 ma). The opposite end of the load line will fall on a point determined by the minimum instantaneous plate voltage, and by the maximum instantaneous plate current. The minimum plate voltage, for best linearity should be considerably higher than Operating Parameters for the Class ABI www.americanradiohistory.com HANDBOOK ABI R.F. Power Amplifiers 167 therefore the load line cannot cross the Eg =0 line. At the point Ep =600, Eg =0, the maximum plate current is 580 ma (Point "B "). Each point the load line crosses a grid voltage axis may be taken as a point for construction of the Eg -Ip curve, just as was done in figure 22, chapter 6. A constructed curve shows that the approximate static bias voltage is -74 volts, which checks closely with point A of figure 14. In actual practice, the bias voltage is set to hold the actual dissipation slightly below the maximum figure of the tube. R F. E our R.F. E iN The single tone power output is: Figure 13 LINEARITY CURVE OF TYPICAL TETRODE AMPLIFIER Emax -Eminx Ipmax. or 4 3000 -600 x .58=348 watts 4 The plate current -angle efficiency factor for this class of operation is 0.73, and the actual At point "A" the instantaneous plate voltage is swinging down to the value of screen voltage. At point "B" it is swinging well below the screen and is approaching the point where saturation, or plate current limiting takes place. plate circuit efficiency is: Np =Emax -Emin x0.73, or Emax 3000- 600 X 0.73 = 58.4% 3000 The power input to the stage is therefore the screen voltage. In this case, the screen voltage is 500, so the minimum plate voltage excursion should be limited to 600 volts. Class AB1 operation implies no grid current, Pox 100 or, 348 Np 58.4 = 595 watts The plate dissipation is: 595 -348 =247 watts. . 4k1RtIoN Pi- network tetrode TOP VIEW OF A 4 -250A AMPLIFIER amplifier may be operated Class AB, Class 8, or Class C by varying potentials applied to tube. Same general physical and mechanical design applies in each cose. www.americanradiohistory.com 168 r, RADIO THE -F Vacuum Tube Amplifiers R oo. 4 -400A rues E SCR = 900 VOLTS r 'S r I0' 4 t5L- t POINT B w o is SO 00 oC woo 2000 ,000 S - ER .s .E .4 m,_ LOAD LINE .2 POINT A o I zs SO I VALUE OF ER MIN OOOV.. 11.2 0.3E A VALUE OF MAX. DISSIPATION (3000 V. X 0.71 A .300 wArrs) 75 2 nni hiclre 14 OPERATING PARAMETERS FOR TETRODE LINEAR AMPLIFIER ARE OBTAINED FROM CONSTANT- CURRENT CURVES. It can be seen that the limiting factor for this class of operation is the static plate dissipation, which is quite a bit higher than the operating dissipation level. It is possible, at the expense of a higher level of distortion, to drop the static plate dissipation and to increase the screen voltage to obtain greater power output. If the screen voltage is set at 800, and the bias increased sufficiently to drop the static plate current to 90 ma, the single tone d-c plate current may rise to 300 ma, for a power input of 900 watts. The plate circuit efficiency is 55.6%, and the power output is 500 watts. Static plate dissipation is 270 watts. At a screen potential of 500 volts, the maximum screen current is less than 1 ma, and under certain loading conditions may be negative. When the screen potential is raised to 800 volts maximum screen current is 18 ma. The performance of the tube depends upon the voltage fields set up within the tube by the cathode, control grid, screen grid, and plate. The quantity of current flowing in the screen circuit is only incidental to the fact that the screen is maintained at a positive potential with respect to the electron stream surrounding it. The tube will perform as expected so long as the screen current, in either direction, does not create undesirable changes in the screen voltage, or cause excessive screen dissipation. Good regulation of the screen supply is there- required. Screen dissipation is highly responsive to plate loading conditions, and the plate circuit should always be adjusted so as to keep the screen current below the maximum dissipation level as established by the applied voltage. G -G Class B Linear Certain tetrode and pentode tubes, such as the 6AG7, Tetrode Amplifier 837, and 803 perform well as grounded grid class B linear amplifiers. In this configuration both grids and the suppressor are grounded, and excitation is applied to the cathode circuit of the tube. So connected, the tubes take on characteristics of high -mu triodes. No bias or screen supplies are required for this type of operation, and reasonably linear fore 6AG7 -r DRIVER B4-300-700 Figure 15 SIMPLE GROUNDED -GRID www.americanradiohistory.com LINEAR AMPLIFIER v. 169 HANDBOOK V2 VI 837 837 500 803 500 803 500 250 250 250 250 VS V4 V3 837 INPUT 2 War TS PEAK .001 2 N = .001 5KV = Ti E0 115V Figure 3 16 -STAGE KILOWATT LINEAR AMPLIFIER FOR 80 OR 40 METER OPERATION operation can be had with a very minimum of circuit components (figure 15). The input impedance of the g -g stage falls between 100 and 250 ohms, eliminating the necessity of swamping resistors, even though considerable power is drawn by the cathode circuit of the g -g stage. Power gain of a g -g stage varies from approximately 20 when tubes of the 6AG7 type are used, down to five or six for the 837 and 803 tubes. One or more g -g stages may be cascaded to provide up to a kilowatt of power, as illustrated in figure 16. The input and output circuits of cascaded g -g stages are in series, and a variation in load impedance of the output stage reflects back as a proportional change on the input circuit. If the first g -g stage is driven by a high impedance source, such as a tetrode amplifier, any change in gain will automatically be compensated for. If the gain of V4 -V5 drops, the input impedance to that stage will rise. This change will reflect through V2 -V3 so that the load impedance of VI rises. Since V1 has a high internal impedance the output voltage will rise when the load impedance rises. The increased output voltage will raise the output voltage of each g -g stage so that the overall output is nearly up to the initial value before the drop in gain of V4 -V5. The tank circuits, therefore, of all g -g stages must be resonated with low plate voltage and excitation applied to the tubes. Tuning of one stage will affect the ocher stages, and the input and coupling of each stage must be adjusted in turn until the proper power limit is reached. Operating Dota for 4 -400A Grounded Grid Linear Amplifier ACH ti 2500 An open frame filament transformer may be used for TI. Cathode taps are ad- justed for proper excitation of following stage. using a 4 -400A tube for the h.f. region. The operating characteristics of the amplifier are summarized in figure 17. It can be noted that unusually low screen voltage is used on the tube. The use of lower screen voltage has the adverse effect of increasing the driving power, but at the same time the static plate current of the stage is decreased and linearity is imimproved. For grounded grid operation of the 4 -400A, a screen voltage of 300 volts (filament to screen) gives a reasonable compromise between these factors. OPERATING DATA FOR 4- 400A/4 -250A G -G. LINEAR AB, AMPLIFIER (SINGLE CONE) D -C SCREEN VOLTAGE D -C PLATE VOLTAGE +300 +3000 +3500 60 MA. STATIC PLATE CURRENT D -C +300 -60 GRID BIAS 67 PEAK CATHODE SWING V. V. 60 MA. -59 V. 113 V. MINIMUM PLATE VOLTAGE 660 MAXIMUM SIGNAL GRID CURRENT 3.6 MA. 10 MAXIMUM SIGNAL SCREEN CURRENT .1 MA. 20 MA. MAXIMUM SIGNAL PLATE CURRENT 195 MA. 267 MA. MAXIMUM SIGNAL PLATE DISSIPATION 235 235 STATIC PLATE DISSIPATION 160 W. 210W. GRID DRIVING POWER 0.63 3.4 FEEDTHRU POWER 6.55 W. ,s.e V. W. W. S00 V. MA. W. W. W. POWER OUTPUT (MAXIMUM) 350 W. 700 W. POWER INPUT (MAX /MUM) 565 W. 935 W. Experiments have been conducted by Collins Radio Co. on a grounded grid linear amplifier stage www.americanradiohistory.com Figure 17 CHAPTER NINE The Oscilloscope The cathode -ray oscilloscope (also called oscillograph) is an instrument which permits visual examination of various electrical phenomena of interest to the electronic engineer. Instantaneous changes in voltage, current and phase are observable if they take place slowly enough for the eye to follow, or if they are periodic for a long enough time so that the eye can obtain an impression from the screen of the cathode -ray tube. In addition, the cathode ray oscilloscope may be used to study any variable (within the limits of its frequency response characteristic) which can be converted into electrical potentials. This conversion is made possible by the use of some type of transducer, such as a vibration pickup unit, pressure pickup unit, photoelectric cell, microphone, or a variable impedance. The use of such a transducer makes the oscilloscope a valuable tool in fields other than electronics. the recipient of signals from two sources: the vertical and horizontal amplifiers. The operation of the cathode -ray tube itself has been covered in Chapter 4; the auxiliary circuits pertaining to the cathode -ray tube will be covered here. The Vertical The incoming signal which is to be examined is applied to the terminals marked Vertical Input and Ground. The Vertical Input terminal is connected through capacitor CE (figure 2) so that the a-c component of the input signal appears across the vertical amplifier gain control potentiometer, R,. Thus the magnitude of the incoming signal may be controlled to provide the desired deflection on the screen of Amplifier To 'ERTICAL NF T 9 -1 A Typical Cathode -Ray Oscilloscope FLIP. INTENSITY MOD. ATE ro HORIZONTAL - -So -- DEFLECTION EAT. T, RE GND For the purpose of analysis, the operation of a simple oscilloscope will be described. The Du Mont type 274-A unit is a fit instrument for such a description. The block diagram of the 274 -A is shown in figure 1. The electron beam of the cathode -ray tube can be moved O -- -rsa - - - - / NoNIZONTAL/ SWEEPS IUT / R-R}DIR7 111111000111111 AM. vertically or horizontally, or the vertical and - DIR 1NFVT Figure BLOCK DIAGRAM, TYPE 274 -A CATHODE -RAY OSCILLOSCOPE horizontal movements may be combined to produce composite patterns on the tube screen. As shown in figure 1, the cathode -ray tube is 1 170 www.americanradiohistory.com SI STNC. Time Base Generator 171 CI 0.25 LF INPUT 6AC7 O--{ OUTPUT VERT. AMP. Ra CONTROL 11A RT.15aK R30 ee M Cz R2 5100 1K Figure 2 TYPICAL AMPLIFIER SCHEMATIC Figure the cathode -ray tube. Also, as shown in figure 1, S, has been incorporated to by -pass the vertical amplifier and capacitively couple the input signal directly to the vertical deflection plate if so desired. In figure 2, V, is a 6AC7 pentode tube which is used as the vertical amplifier. As the signal variations appear on the grid of variations in the plate current of V, will take place. Thus signal variations will appear in opposite phase and greatly amplified across the plate resistor, R,. Capacitor C, has been added across R, in the cathode circuit of V, to flatten the frequency response of the amplifier at the high frequencies. This capacitor because of its low value has very little effect at low input frequencies, but operates more effectively as the frequency of the signal increases. The amplified signal delivered by V, is now applied through the second half of switch S, and capacitor C, to the free vertical deflection plate of the cathode -ray tube (figure 3). V The circuit of the horizontal amplifier and the circuit of the vertical amplifier, described in the above paragraph, are similar. A switch in the input circuit makes provision for the input from the Horizontal Input terminals to be capacitively coupled to the grid of the horizontal amplifier or to the free horizontal deflection plate thus by- passing the amplifier, or for the output of the sweep generator to be capacitively coupled to the amplifier, as shown in figure 1. The Horizontal Amplifier The Time Bose Generator e l e c t r i c al wave forms by the use of a cathode -ray tube frequently 3 SCHEMATIC OF CATHODE -RAY TUBE CIRCUITS A 5BPIA cathode -ray tube is used in this instrument. As shown, the necessary potentials for operating this tube are obtained from a voltage divider mode up of resistors R21 through R26 inclusive. The intensity of the beam is adjusted by moving the contact on R21. This adjusts the potential on the cathode more or less negative with respect to the grid which is operated at the full negative voltage -1200 volts. Focusing to the desired sharpness is accomplished by adjusting the contact on R23 to provide the correct potential for anode no. 1. Interdependency between the focus and the, intensity controls is inherent in all electrostatically focused cathode-ray tubes. In short, there is an optimum setting of the focus control for every setting of the intensity control. The second anode of the 5BP IA is operated at ground potential in this instrument. Also one of each pair of deflection plates is operated at ground potential. The cathode is operated at a high negative potential (approximately 1200 volts) so that the total overall accelerating voltage of this tube is regarded os 1200 volts since the second anode is operated at ground potential. The vertical and horizontal positioning controls which are connected to their respective deflection plates are capable of supplying either a positive or negative d -c potential to the deflection plates. This permits the spot to be positioned at any desired place on the entire screen. Investigation of requires that some means be readily available to determine the variation in these wave forms with respect to time. When such a time base is required, the patterns presented on the cathode -ray tube screen show the variation in amplitude of the input signal with respect to time. Such an arrangement is made possible by the inclusion in the oscilloscope of a Time Base- Generator. The function of this generator is to move the spot across the screen at a constant rate from left to right between two selected points, to return the spot almost instantaneously to its original position, and to www.americanradiohistory.com 172 The Oscilloscope 884 tube, the tube THE RADIO will ionize (or fire) at a specific plate voltage. Capacitors C3.-C24 are selectively connected in parallel with the 884 tube. Resistor limits the peak current drain of the gas triode. The plate voltage on this tube is obtained through resistors R22, and R11. The voltage applied to the plate of the 884 tube cannot reach the power supply voltage because of the charging effect this voltage has upon the capacitor which is connected across the tube. This capacitor charges until the plate voltage becomes high enough to ionize the gas in the tube. At this time, the 884 tube starts to conduct and the capacitor discharges through the tube until its voltage falls to the extinction potential of the tube. When the tube stops conducting, the capacitor voltage builds up until the tube fires again. As this action continues, it results in the sawtooth wave form of figure 4 appearing at the junction of and R77. R Figure SAWTOOTH R 4 WAVE FORM repeat this procedure at a specified rate. This action is accomplished by the voltage output from the time base (sweep) generator. The rate at which this voltage repeats the cycle of sweeping the spot across the screen is referred to as the sweep frequency. The sweep voltage necessary to produce the motion described above must be of a sawtooth waveform, such as that shown in figure 4. The sweep occurs as the voltage varies from A to B, and the return trace as the voltage varies from B to C. If A-B is a straight line, the sweep generated by this voltage will be linear. It should be realized that the sawtooth sweep signal is only used to plot variations in the vertical axis signal with respect to time. Specialized studies have made necessary the use of sweep signals of various shapes which are introduced from an external source through the Horizontal Input terminals. R Synchronization the sweep generator may be synchronized from the vertical amplifier or from an external source. The switch S, shown in figure 5 is mounted on the front panel to be easily accessible to the operator. If no synchronizing voltage is applied, the discharge tube will begin to conduct when the plate potential reaches the value of F.t (Firing Potential). When this breakdown takes place and the tube begins to conduct, the capacitor is discharged rapidly through the tube, and the plate voltage decreases until it reaches the extinction potential E1. At this point conduction ceases, and the plate potential rises slowly as the capacitor begins to charge through R7, and R25. The plate potential will again reach a point of conduction and the circuit will start a new cycle. The rapidity of the plate voltage rise is dependent upon the circuit constants R77 R25, and the capacitor selected, The sawtooth voltage necessary to obtain the linear time base is generated by the circuit of figure 5, which operates as follows: A type 884 gas triode (V3) is used for the sweep generator tube. This tube contains an inert gas which ionizes when the voltage between the cathode and the plate reaches a certain value. The ionizing voltage depends upon the bias voltage of the tube, which is determined by the voltage divider resistors R12 -R17. With a specific negative bias applied to the The Sawtooth Generator - C10,0 5 ur. GÉNECN1T011 CII,O.IUr C 12, 03 C T OUTPUT ur 1001212! CN,22012121 SIGNAL TO DEFLECTION PLATE Rr ISO 470 RE 6AC7 E Provision has been made so 0 nTERr.A Ree Su EAT. FINE MM CONTROL SYNC R a100A 1200 Figure SCHEMATIC OF TO St 5 SWEEP GENERATOR www.americanradiohistory.com HANDBOOK The Oscilloscope 173 Eb + EPVSEg EP+ STATIC CONTROL f CHARACTERISTIC Er - -- Irl I FREE RUN HING PERIOD t+ D.C.GRID BIAS FIRING POTENTIAL (D.C. BIAS) Eex FIRING 011 POTENTIA WITH SYNC. SIGNAL SYNCHRONIZE PERIOD EXTINCTION POTENTIAL -EJ SYNC. SIGNAL APPLIED TO GRID Figure 6 ANALYSIS OF SYNCHRONIZATION OF C10 -C14, as well as the supply voltage The exact relationship is given by: Ec Eb(1_erct Eb TIME -BASE GENERATOR time the plate potential rises to a sufficient value, so that the sweep recurs at the same or an integral sub-multiple of the synchronizing signal rate. This is illustrated in figure 6. ) Figure - shows the power supply to be made up of two definite sections: a low voltage positive supply which provides power for operating the amplifiers, the sweep generator, and the positioning circuits of the cathode -ray tube; and the high voltage negative supply which provides the potentials necessary for operating the various Power Supply Where E,----=Capacitor voltage at time t Eb =Supply voltage (B+ supply - cathode bias) Er---Firing potential or potential at which time-base gas triode fires Ex =Extinction potential or potential at which time-base gas triode ceases to conduct e= Base of natural logarithms t= Time in seconds r =Resistance c =Capacity in in ohms (R=T + Rzs) farads (C10, II, 12, ,,, or 14) The frequency of oscillation will be approximately: 1 f=rc Et-Ex Ebl Under this condition (no synchronizing signal applied) the oscillator is said to be /ree running. When a positive synchronizing voltage is applied to the grid, the firing potential of the tube is reduced. The tube therefore ionizes at a lower plate potential than when no grid signal is applied. Thus the applied snychronizing voltage fires the gas -filled triode each Figure 7 SCHEMATIC OF POWER SUPPLY www.americanradiohistory.com 174 The Oscilloscope THE RADIO I 0001 ono o p o .^ .52.40) IÓ .rIN 410 o co > U \r > Y o } 7 t N xn ¢ 0 H r = JV' úu ÿJ Uà VIQ I 1\ á ñl ai U nf z o I www.americanradiohistory.com HANDBOOK TIME Display of Waveforms 175 -+ 4 SEC. IFigure 9 PROJECTION DRAWING OF A SINEWAVE APPLIED TO THE VERTICAL AXIS AND A SAWTOOTH WAVE OF THE SAME FRE- QUENCY APPLIED SIMULTANEOUSLY THE HORIZONTAL AXIS ON electrodes of the cathode -ray tube, and for certain positioning controls. The positive low voltage supply consists of full -wave rectifier (V,), the output of which is filtered by a capacitor input filter (20 -20 µfd. and 8 II). It furnishes approximately 400 volts. The high voltage power supply employs a half wave rectifier tube, V,. The output of this rectifier is filtered by a resistance -capacitor filter consisting of 0.5 -0.5 pfd. and .18 M. A voltage divider network attached from the output of this filter obtains the proper operating potentials for the various electrodes of the cathode -ray tube. The complete schematic of the Du Mont 274 -A Oscilloscope is shown in figure 8. 9 -2 Display of Waveforms Together with a working knowledge of the controls of the oscilloscope, an understanding of how the patterns are traced on the screen must be obtained for a thorough knowledge of oscilloscope operation. With this in mind a careful analysis of two fundamental waveform patterns is discussed under the following headings: a. Patterns plotted against time (using the sweep generator for horizontal deflection). b. Lissajous Figures (using a sine wave for horizontal deflection). Patterns Plotted Against Time A sine wave is typical of such a pattern and is con- venient for this study. This Figure 10 PROJECTION DRAWING SHOWING THE RESULTANT PATTERN WHEN THE FREQUENCY OF THE SAWTOOTH IS ONE -HALF OF THAT EMPLOYED IN FIGURE 9 amplified by the vertical amplifier and impressed on the vertical (Y -axis) deflec- wave is tion plates of the cathode -ray tube. Simultaneously the sawtooth wave from the time base generator is amplified and impressed on the horizontal (X -axis) deflection plates. The electron beam moves in accordance with the resultant of the sine and sawtooth signals. The effect is shown in figure 9 where the sine and sawtooth waves are graphically represented on time and voltage axes. Points on the two waves that occur simultaneously are numbered similarly. For example, point 2 on the sine wave and point 2 on the sawtooth wave occur at the same instant. Therefore the position of the beam at instant 2 is the resultant of the voltages on the horizontal and vertical deflection plates at instant 2. Referring to figure 9, by projecting lines from the two point 2 positions, the position of the electron beam at instant 2 can be located. If projections were drawn from every other instantaneous position of each wave to intersect on the circle representing the tube screen, the intersections of similarly timed projections would trace out a sine wave. In summation, figure 9 illustrates the principles involved in producing a sine wave trace on the screen of a cathode -ray tube. Each intersection of similarly timed projections represents the position of the electron beam acting under the influence of the varying voltage waveforms on each pair of deflection plates. Figure 10 shows the effect on the pattern of decreasing the frequency of the sawtooth www.americanradiohistory.com 176 The Oscilloscope THE RADIO B Figure 12 METHOD OF CALCULATING FREQUENCY RATIO OF LISSAJOUS FIGURES Figure 11 PROJECTION DRAWING SHOWING THE RE-SULTANT LISSAJOUS PATTERN WHEN A SINE WAVE APPLIED TO THE HORIZONTAL AXIS IS THREE TIMES THAT APPLIED TO THE VERTICAL AXIS wave. Any recurrent waveform plotted against time can be displayed and analyzed by the same procedure as used in these examples. The sine wave problem just illustrated is typical of the method by which any waveform can be displayed on the screen of the cathode ray tube. Such waveforms as square wave, sawtooth wave, and many more irregular recurrent waveforms can be observed by the same method explained in the preceding paragraphs. 9 -3 Obtaining a Lissalous Pattern on the screen Oscilloscope Settings 1. The horizontal am- plifier should be discon- nected from the sweep oscillator. The signal to be examined should be connected to the horizontal amplifier of the oscilloscope. 2. An audio oscillator signal should be connected to the vertical amplifier of the oscilloscope. 3. By adjusting the frequency of the audio oscillator a stationary pattern should be obtained on the screen of the oscilloscope. It is not necessary to stop the pattern, but merely to slow it up enough to count the loops at the side of the pattern. 4. Count the number of loops which intersect an imaginary vertical line AB and the number of loops which intersect the imaginary horizontal line BC as in figure 12. The ratio of the number of loops which intersect AB is to Lissajous Figures Another fundamental pattern is the Lissajous figure, named after the 19th century French scientist. This type of pattern is of particular use in determining the frequency ratio between two sine wave signals. If one of these signals is known, the other can be easily calculated from the pattern made by the two signals upon the screen of the cathode -ray tube. Common practice is to connect the known signal to the horizontal channel and the unknown signal to the vertical channel. The presentation of Lissajous figures can be analyzed by the same method as previously used for sine wave presentation. A simple example is shown in figure 11. The frequency ratio of the signal on the horizontal axis to the signal on the vertical axis is 3 to 1. If the known signal on the horizontal axis is 60 cycles per second, the signal on the vertical axis is 20 cycles. O RATIO I O I RATIO O RATIO 5 Figure 13 OTHER LISSAJOUS PATTERNS www.americanradiohistory.com 2I HANDBOOK HHASE Lissajous DIFFERENCE =O PHASE DIFFERENCE 190 PHASE DIFFERENCE PHASE -5 OIFFERENCE'225 Figure LISSAJOUS PATTERNS OBTAINED PHASE DIFFERENCE Figure 13 shows other examples of Lissa jous figures. In each case the frequency ratio shown is the frequency ratio of the signal on the horizontal axis to that on the vertical ve r ti c axis. Phase Differonce Patterns Coming under the heading of Lissajous figures is the method used to determine the phase difference between signals of the same frequency. The patterns i n vol v e d take on the form of ellipses with different degrees of ec- centricity. The following steps should be taken to obphase -difference pattern: 1. With no signal input to the oscilloscope, the spot should be centered on the screen of the tube. 2. Connect one signal to the vertical amplifier of the oscilloscope, and the other signal to the horizontal amplifier. 3. Connect a common ground between the two frequencies under investigation and the oscilloscope. .90. 270 177 PHASE DIFFERENCE=135 PHASE DIFFERENCE 315 14 FROM THE the number of loops which intersect BC as the frequency of the horizontal signal is to the frequency of the vertical signal. tain PHASE DIFFERENCE Figures MAJOR PHASE DIFFERENCE ANGLES plifier control is adjusted (3 inches). Reconnect the signal to the vertical amplifier. The resulting pattern will give an accurate picture of the exact phase difference between the two waves. If these two patterns are exactly the same frequency but different in phase and maintain that difference, the pattern on the screen will remain stationary. If, 'however, one of these frequencies is drifting slightly, the pattern will drift slowly through 360°. The phase angles of 0 °, 45 °, 90 °, 135 °, 180 °, 225 °, 270 °, 315° are shown in figure 14. Each of the eight patterns in figure 14 can be analyzed separately by the previously used a Adjust the vertical amplifier gain so as to give about 3 inches of deflection on a 5 inch tube, and adjust the calibrate d scale of the oscilloscope so that the vertical axis of the scale coincides precisely with the vertical deflection of the spot. 5. Remove the signal from the vertical amplifier, being careful not to change the setting of the vertical gain control. 6. Increase the gain of the horizontal amplifier to give a deflection exactly the same as that to which the vertical am- TIME - 4. Figure 15 PROJECTION DRAWING SHOWING THE RESULTANT PHASE DIFFERENCE PATTERN OF TWO SINE WAVES 45° OUT OF PHASE www.americanradiohistory.com 178 Oscilloscope The Y Y INTERCEPT =O / SINE Y THE MAXIMUM =I Y MAXIMUM. SINE= MAXIMUM = I 'I MA IMU MAXIMUM= I SINE e= _S So INTERCEPT'.S SINEe', s=150- projecti ',n method. Figure 15 shows two sine waves which differ in phase being projected on to the screen of the cathode -ray tube. These signals represent a phase difference of 45 °. It is extremely important: (1) that the spot has been centered on the screen of the cathode ray tube, (2) that both the horizontal and vertical amplifiers have been adjusted to give exactly the same gain, and (3) that the calibrated scale be originally set to coincide with the displacement of the signal along the vertical axis. If the amplifiers of the oscilloscope are not used for conveying the signal to the deflection plates of the cathode -ray tube, the coarse frequency switch should be set to horizontal input direct and the vertical input MODULATED Y 'I DIFFERENCE Figure 17 MODULATION I ¿;-:. Figure 16 EXAMPLES SHOWING THE USE OF THE FORMULA TRAPEZOIDAL = INTERCEPT =.5 s' o YINTERCEil, s=so Y RADIO FOR DETERMINATION OF PHASE switch to direct and the outputs of the two signals must be adjusted to result in exactly the same vertical deflection as horizontal deflection. Once this deflection has been set by either the oscillator output controls or the amplifier gain controls in the oscillograph, it should not be changed for the duration of the measurement. Determination of the Phase Angle The relation commonly used in determining the phase angle between signals is: Y intercept Sine 9 Y maximum PATTERN Figure 18 CARRIER WAVE PATTERN Figure 19 PROJECTION DRAWING SHOWING TRAPEZOIDAL PATTERN www.americanradiohistory.com HANDBOOK Trapezoidal Pattern MODULATED CARRIER R F. 179 POWER AMPLIFIER 1-0 ANTENNA TIME EACH 1M, 1 MODULATOR 500.11ÁF STAGE - /SAW TOOTH 10000 V. TV CAPACITOR SWEEP CRO LC TUNES TO OPERATING FREQUENCY Figure 20 PROJECTION DRAWING SHOWING MODOLATED CARRIER WAVE PATTERN C e+ intercept = = Y maximum = where Y 9 phase angle between signals point where ellipse crosses vertical axis measured in tenths of inches. (Calibrations on the calibrated screen) highest vertical point on ellipse in tenths of inches Several examples of the use of the formula are given in figure 16. In each case the Y intercept and Y maximum are indicated together with the sine of the angle and the angle itself. For the operator to observe these various patterns with a single signal source such as the test signal, there are many types of phase shifters which can be used. Circuits can be obtained from a number of radio text books. The procedure is to connect the original signal to the horizontal channel of the oscilloscope and the signal which has passed through the phase shifter to the vertical channel of the oscilloscope, and follow the procedure set forth in this discussion to observe the various phase shift patterns. 9 -4 Monitoring Transmitter Performance with the Oscilloscope The oscilloscope may be used as an aid for the proper operation of a radiotelephone transmitter, and may be used as an indicator of the overall performance of the transmitter output signal, and as a modulation monitor. There are two types of patterns that can serve as indicators, the trapezoidal pattern (figure 17) and the modu- Waveforms NOTE' IF L _ PICKUP IS INSUFFICIENT, A TUNED CIRCUIT MAY BE USED AT THE OSCILLOSCOPE AS SHOWN. R F. Figure 21 MONITORING CIRCUIT FOR TRAPEZOIDAL MODULATION PATTERN laced wave pattern (figure 18). The trapezoidal pattern is presented on the screen by impressing a modulated carrier wave signal on the vertical deflection plates and the s i g n a l t h a t modulates the carrier wave signal (the modulating signal) on the horizontal deflection plates. The trapezoidal pattern can be analyzed by the method used previously in analyzing waveforms. Figure 19 shows how the signals cause the electron beam to trace out the pattern. The modulated wave pattern is accomplished by presenting a modulated carrier wave on the vertical deflection plates and by using the time -base generator for horizontal deflection. The modulated wave pattern also can be used for analyzing waveforms. Figure 20 shows how the two signals cause the electron beam to trace out the pattern. oscilloscope connections for obtaining a trapezoidal pattern are shown in figure 21. A portion of the audio output of the transmitter modulator is applied to the horizontal input of the oscilloscope. The vertical amplifier of the oscilloscope is disconnected, and a small amount of modulated r -f energy is coupled directly to the vertical d e f l e c t i o n plates of the oscilloscope. A small pickup loop, loosely coupled to the final amplifier tank circuit and connected to the vertical deThe Trapezoidal Pattern www.americanradiohistory.com The Oscilloscope The 180 T H E RADIO i T EMIN E MAX 1 TRAPEZOIDAL WAVE PATTERN Figure 22 Figure (L ESS THAN 100^; MODULATION) (100 mula: Emax Emax = - Emin x t Emin Figure 24 MODULATION) flection plates by a short length of coaxial line will suffice. The amount of excitation to the plates of the oscilloscope may be adjusted to provide a pattern of convenient size. Upon modulation of the transmitter, the trapezoidal pattern will appear. By changing the degree of modulation of the carrier wave the shape of the pattern will change. Figures 22 and 23 show the trapezoidal pattern for various degrees of modulation. The percentage of modulation may be determined by the following forModulation percentage 23 (OVER MODULATION) figure 25. The internal sweep circuit of the oscilloscope is applied to the horizontal plates, and the modulated r -f signal is applied to the vertical plates, as described before. If desired, the internal sweep circuit may be snychronized with the modulating signal of the transmitter by applying a small portion of the modulator output signal to the external sync post of the oscilloscope. The percentage of modulation may be determined in the same fashion as with a trapezoidal pattern. Figures 26, 27 and 28 show the modulated wave pattern for various degrees of modulation. 100 where Emax and Emin are defined as in figure 22. An overmodulated signal is shown in figure 9 -5 Receiver -F Alignment with an Oscilloscope I 24. The Modulated Wove Pattern R F. The oscilloscope connections for obtaining a modulated wave pattern are shown in POWER AMPLIFIER TO CRO ANTENNA USE INTERNAL SWEE a FROM MODUL ATOR LC TUNES TO OP- ERATING FREQUENCY Figure 25 MONITORING CIRCUIT FOR MODULATED WAVE PATTERN The alignment of the i -f amplifiers of a receiver consists of adjusting all the tuned circuits to resonance at the intermediate frequency and at the same time to permit passage of a predetermined number of side bands. The best indication of this adjustment is a resonance curve representing the response of the i -f circuit to its particular range of frequencies. As a rule medium and low- priced receivers use i -f transformers whose bandwidth is about 5 kc. on each side of the fundamental frequency. The response curve of these i -f transformers is shown in figure 29. High fidelity receivers usually contain i -f transformers which have a broader bandwidth which is usually 10 kc. on each side of the fundamental. The response curve for this type transformer is shown in figure 30. Resonance curves such as these can be displayed on the screen of an oscilloscope. For a complete understanding of the procedure it is important to know how the resonance curve is traced. www.americanradiohistory.com HANDBOOK Receiver Alignment VV EMIN E 181 V\I" MLY. 1\1\ CARRIER WAVE PATTERN Figure 27 Figure 26 (100% MODULATION) (LESS THAN 100% MODULATION) The Resonance Curve on the Screen To present a resonance curve on the screen, a frequency modulated signal source must be avail a b l e. Figure 28 This signal source is a signal generator whose output is the fundamental i -f frequency which is frequency- modulated 5 to 10 kc. each side of the fundamental frequency. A signal generator of this type generally takes the form of an ordinary signal generator with a rotating motor driven tuned circuit capacitor, called a uwob- (OVER MODULATION) bulator, or its electronic equivalent, a react- ance tube. The method of presenting a resonance curve on the screen is to connect the vertical channel of the oscilloscope across the detector load of the receiver as shown in the detectors of figure 31 (between point A and ground) and the time -base generator output to the horizontal channel. In this way the d -c voltage across the detector load varies with the frequencies which are passed by the i -f system. Thus, if the time -base generator is set at the frequency of rotation of the motor driven capacitor, or the reactance tube, a pattern resembling figure 32, a double resonance curve, appears on the screen. Figure 32 is explained by considering fighalf a rotation of the motor driven capacitor the frequency increases from 445 kc. to 465 kc., more than covering the range of frequencies passed by the i -f system. Therefore, a full resonance curve is presented on the screen during this half cycle of rotation since only half a cycle of the voltage producing horizontal deflection has transpired. In the second half of the rotation the motor ure 33. In eKC K 4 KC ecc Figure 29 FREQUENCY RESPONSE CURVE OF THE I -F OF A LOW PRICED RECEIVER ecc TRIODE DETECTOR eKc DIODE DETECTOR Figure Figure 30 FREQUENCY RESPONSE OF HIGH- FIDELITY I -F SYSTEM 31 CONNECTION OF THE OSCILLOSCOPE ACROSS THE DETECTOR LOAD www.americanradiohistory.com The 182 Figure 32 DOUBLE RESONANCE CURVE 445 KC 455 KC THE Oscilloscope 46 455 KC KC 445 KC Figure 33 DOUBLE RESONANCE ACHIEVED BY COMPLETE ROTATION OF THE MOTOR DRIVEN CAPACITOR curve is observed as it sweeps the spot across the screen from left to right; and it is observed again as the sine wave sweeps the spot back again from right to left. Under these conditions the two response curves are superimposed on each other and the high frequency responses of both curves are at one end and the low frequency response of both curves is at the other end. The i -f trimmer capacitors are adjusted to produce a response curve which is symmetrical on each side of the fundamental frequency. When using sawtooth sweep, the two response curves can also be superimposed. If the sawtooth signal is generated at exactly twice the frequency of rotation of the motor driven capacitor, the two resonance curves will be superimposed (figure 34) if the i -f transformers are properly tuned. If the two curves do not coincide the i -f trimmer capacitors should be adjusted. At the point of coincidence the tuning is correct. It should be pointed out that rarely do the two curves agree perfectly. As a result, optimum adjustment is made by making the peaks coincide. This latter procedure is the one generally used in i -f adjustment. When the two curves coincide, it is evident that the i -f system responds equally to signals higher and lower than the fundamental i -f frequency. 9 -6 Figure 34 SUPER -POSITION OF RESONANCE CURVES driven capacitor takes the frequency of the signal in the reverse order through the range of frequencies passed by the i -f system. In this interval the time -base generator sawtooth waveform completes its cycle, drawing the electron beam further across the screen and then returning it to the starting point. Subsequent cycles of the motor driven capacitor and the sawtooth voltage merely retrace the same pattern. Since the signal being viewed is applied through the vertical amplifier, the sweep can be synchronized internally. Some signal generators, particularly those employing a reactance tube, provide a sweep output in the form of a sine wave which is synchronized to the frequency with which the reactance tube is swinging the fundamental frequency through its limits, usually 60 cycles per second. If such a signal is used for horizontal deflection, it is already synchronized. Since this signal is a sine wave, the response RADIO Single Sideband Applications Measurement of power output and distortion are of particular importance in SSB transmitter adjustment. These measurements are related to the extent that distortion rises rapidly when the power amplifier is overloaded. The useable power output of a SSB transmitter is often defined as the maximum peak envelope power II11IIIIIIU114ulul m IIIIIIIIIIIIIIIIIIII Figure SINGLE 35 TONE PRESENTATION Oscilloscope trace of SSB signal modulated by single tone (A). Incomplete carrier supression or spurious products will show modulated envelope of (B). The ratio of supression is: www.americanradiohistory.com S - 20 log A +B A -B S.S.B. HANDBOOK R POWER VER A -F INPUT T R -F LIFIER SSB INPUT VOLTAGE FOM TEST GERMANIUM DIODE tpplications 2.5 183 MM RFC AUDIO OUTPUT 70 OSCILLOSCOPE DIVIDER OR PICMUP COIL INPUT ENVELOPE DETECTOR Figure 37 SCHEMATIC OF ENVELOPE DETECTOR OSCILLOSCOPE Figure 36 BLOCK DIAGRAM OF LINEARITY TRACER obtainable with a specified signal-to- distortion ratio. The oscilloscope is a useful instrument for measuring and studying distortion of all types that may be generated in single sideband equipment. Single Tone When aSSB transmitter is modu- laced with a single audio tone, the r -f output should be a single radio frequency. If the vertical plates of the oscilloscope are coupled to the output of the transmitter, and the horizontal amplifier sweep is set to a slow rate, the scope presentation will be as shown in figure 35. If unwanted distortion products or carrier are present, the top and bottom of the pattern will develop a "ripple" proportional to the degree of spurious Observations products. The linearity tracer is an auxiliary detector to be used with an oscilloscope for quick observation of amplifier adjustments and parameter variations. This instrument consists of two SSB envelope detectors the outputs of which connect to the horizontal and vertical inputs of an oscilloscope. Figure 36 shows a block diagram of atypical linearity test set -up. A two -tone test signal is normally employed to supply a SSB modulation envelope, but any modulating signal that provides an envelope that varies from zero to full amplitude may be The Linearity Tracer used. Speech modulation gives a satisfactory trace, so that this instrument may be used as a visual monitor of transmitter linearity. It is particularly useful for monitoring the signal level and clearly shows when the amplifier under observation is overloaded. The linearity trace will be a straight line regardless of the envelope shape if the amplifier has no distortion. Overloading causes a sharp break in the linearity curve. Distortion due to too much bias is also easily observed and the adjustment for low distortion can easily be made. Another feature of the linearity detector is that the distortion of each individual stage can be observed. This is helpful in troubleshooting. By connecting the input envelope detector to the output of the SSB generator, the overall distortion of the entire r -f circuit beyond this point is observed. The unit can also serve as a voltage indicator which is useful in making tuning adjustments. The circuit of a typical envelope detector is shown in figure 37. Two matched germainum diodes are used as detectors. The detectors are not linear at low signal levels, but if the nonlinearity of the two detectors is matched, the effect of their nonlinearity on the oscilloscope trace is cancelled. The effect of diode differences is minimized by using a diode load of 5,000 to 10,000 ohms, as shown. It is important that both detectors operate at approximately the same signal level so that their differences will cancel more exactly. The operating level should be 1 -volt or higher. It is convenient to build the detector in a small shielded enclosure such as an i -f transformer can fitted with coaxial input and output connectors. Voltage dividers can be similarly constructed so that it is easy to insert the desired amount of voltage attenuation from the various sources. In some cases it is convenient to use a pickup loop on the end of a short length of coaxial cable. The phase shift of the amplifiers in the oscilloscope should be the same and their frequency response should be flat out to at least twenty times the frequency difference of the two test tones. Excellent high frequency characteristics are necessary because the rectified SSB envelope contains harmonics extending to the limit of the envelope detector's response. Inadequate frequency response of the vertical amplifier may cause a little "foot" to appear on the lower end of the trace, as shown in figure 38. If it is small, it may be safely neg- lected. Another spurious effect often encountered is a double trace, as shown in figure 39. This can usually be corrected with an R -C network placed between one detector and the oscilloscope. The best method of testing the detectors and the amplifiers is to connect the input of www.americanradiohistory.com 184 The Oscilloscope OUTPUT SIGNAL LEVEL Figure 38 EFFECT OF INADEQUATE RESPONSE OF VERTICAL AMPLIFIER INPUT SIGNAL LEVEL Figure 41 ORDINATES ON LINEARITY CURVE FOR 3RD ORDER DISTORTION EQUATION Figure 39 DOUBLE TRACE CAUSED BY PHASE SHIFT the envelope detectors in parallel. A perfectly straight line trace will result when everything is working properly. One detector is then connected to the other r -f source through a voltage divider adjusted so that no appreciable change in the setting of the oscilloscope amplifier controls is required. Figure 40 illustrates some typical linearity traces. Trace A is caused by inadequate static plate current in class A or class B amplifiers or a mixer stage. To regain linearity, the grid bias of the stage should be reduced, the screen voltage should be raised, or the signal level should be decreased. Trace B is a result of poor grid circuit regulation when grid current is drawn, or a result of non- linear plate characteristics of the amplifier tube at large plate swings. More grid swamping should be used, or the exciting signal should be reduced. A combination of the effects of A and B are shown in Trace C. Trace D illustrates amplifier overloading. The exciting signal should be reduced. A means of estimating the distortion level observed is quite useful. The first and third order distortion components may be derived by an equation that will give the approximate signal -to- distortion level ratio of a two tone test signal, operating on a given linearity curve. Figure 41 shows a linearity curve with two ordinates erected at half and full peak input signal level. The length of the ordinates et and e2 may be scaled and used in the following equation: Signal -to- distortion ratio in db =20 log 8 e t -e2 TYPICAL LINEARITY TRACES Figure 40 TYPICAL LINEARITY TRACES www.americanradiohistory.com 2 el -e2 CHAPTER TEN Special Vacuum Tube Circuits A whole new concept of vacuum tube applications has been developed in recent years. No longer are vacuum tubes chained to the field of communication. This chapter is devoted to some of the more common circuits encountered in industrial and military applications of the vacuum tube. 10 -1 The characteristics of a diode tube are such that the tube conducts only when the plate is at a positive potential with respect to the cathode. A positive potential may be placed on the cathode, but the tube will not conduct until the voltage on the plate rises above an equally positive value. As the plate becomes more positive with respect to the cathode, the diode conducts and passes that portion of the wave that is more positive than the cathode voltage. Diodes may be used as either series or parallel limiters, as shown in figure 1. A diode may be so biased that only a certain portion of the positive or negative cycle is removed. Diode Limiters Limiting Circuits The term limiting refers to the removal or suppression by electronic means of the extremities of an electronic signal. Circuits which perform this function are referred to as limiters or clippers. Limiters are useful in wave-shaping circuits where it is desirable to square off the extremities of the applied signal. A sine wave may be applied to a limiter circuit to produce a rectangular wave. A peaked wave may be applied to a limiter circuit to eliminate either the positive or negative peaks from the output. Limiter circuits are employed in FM receivers where it is necessary to limit the amplitude of the signal applied to the detector. Limiters may be used to reduce automobile ignition noise in short -wave receivers, or to maintain a high average level of modulation in a transmitter. They may also be used as protective devices to limit input signals to special circùits. An audio peak clipper consisting of two diode limiters may be used to limit the amplitude of an audio signal to a predetermined value to provide a high average level of modulation without danger of overmodulation. An effective limiter for this service is the series -diode gate clipper. A circuit of this clipper is shown in figure 2. The audio signal to be clipped is coupled to the clipper through C,. R, and R2 are the clipper input and output load resistors. The clipper plates are tied together and are connected to the clipping level control, R., through the series resistor, R3. R. acts as a voltage divider between the high voltage supply and ground. The exact point at which clipAudio Peak Limiting 185 www.americanradiohistory.com Special Vacuum Tube 186 e iN Circuits THE RADIO e OUT E IAA' E e IN = VOLTAGE DROP ACROSS DIODE E= VOLTAGE DROP ACROSS DIODE e OUT PIN e OVT E E lTV i VT -A-21 ear Figure e OUT 1 VARIOUS DIODE LIMITING CIRCUITS Series diodes limiting positive and negative peaks are shown in A and ing positive and negative peaks are shown in C and D. Parallel diodes 8. Parallel diodes limitlimiting above and below ground are shown in E and F. Parallel diode limiters which pass negative and positive peaks are shown in G and H. ping will occur is set by R,, which controls the positive potential applied to the diode plates. Under static conditions, a d -c voltage is obtained from R4 and applied through R, to both plates of the 6AL5 tube. Current flows through R,, and divides through the two diode sections of the 6AL5 and the two load resistors, R, and Rr. All parts of the clipper circuit are maintained at a positive potential above ground. The voltage drop between the plate and cathode of each diode is very small compared to the drop across the 300,000 -ohm resistor (R,) in series with the diode plates. The plate and cathode of each diode are therefore maintained at approximately equal potentials as long as there is plate current flow. Clipping does not occur until the peak audio input voltage reaches a value greater than the static voltages at the plates of the diode. R Assume that R4 has been set to a point that will give 4 volts at the plates of the 6AL5. When the peak audio input voltage is less than 4 volts, both halves of the tube conduct at all times. As long as the tube conducts, its resistance is very low compared with the plate resistor R,. Whenever a voltage change occurs across input resistor the voltage at all of the tube elements increases or decreases by the same amount as the input voltage change, and the voltage drop across R, changes by an equal amount. As long as the peak input voltage is less than 4 volts, the 6AL5 acts merely as a conductor, and the output cathode is permitted to follow all voltage changes at the input cathode. If, under static conditions, 4 volts appear at the diode plates, then twice this voltage (8 volts) will appear if one of the diode circuits www.americanradiohistory.com R Clamping Circuits HANDBOOK Ra 6AL5 CLIPPING LEVEL 300K CONTROL C2 Ci 0.1 0.1 e e IN OUT R 00 m R2 R zoom eIN E Bt E 200K Figure 2 THE SERIES -DIODE GATE CLIPPER FOR AUDIO PEAK LIMITING is opened, removing its d -c load from the circuit. As long as only one of the diodes continues to conduct, the voltage at the diode plates cannot rise above twice the voltage selected by R. In this example, the voltage cannot rise above 8 volts. Now, if the input audio voltage applied through C, is increased to any peak value between zero and plus 4 volts, the first cathode of the 6AL5 will increase in voltage by the same amount to the proper value between 4 and 8 volts. The other tube elements will assume the same potential as the first cathode. However, the 6AL5 plates cannot increase more than 4 volts above their original 4 -volt static level. When the input voltage to the first cathode of the 6AL5 increases to more than plus 4 volts, the cathode potential increases to more than 8 volts. Since the plate circuit potential remains at 8 volts, the first diode section ceases to conduct until the input voltage across R, drops below 4 volts. When the input voltage swings in a negative direction, it will subtract from the 4 -volt drop across R, and decrease the voltage on the input cathode by an amount equal to the input voltage. The plates and the output cathode will follow the voltage level at the input cathode as long as the input voltage does not swing below minus 4 volts. If the input voltage does not change more than 4 volts in a negative direction, the plates of the 6AL5 will also become negative. The potential at the output cathode will follow the input cathode voltage and decrease from its normal value of 4 volts until it reaches zero potential. As the input cathode voltage decreases to less than zero, e IVENPOSITIVE A WNEDNGRIDOSEDR Figure 3 LIMITING CIRCUIT GRID the plates will follow. however, the output will stop at zero cathode, grounded through potential as the plate becomes negative. Conduction through the second diode is impossible under these conditions. The output cathode remains at zero potential until the voltage at the input cathode swings back to zero. The voltage developed across output resistor R2 follows the input voltage variations as long as the input voltage does not swing to a peak value greater than the static voltage at the diode plates, determined by R. Effective clipping may thus be obtained at any desired R level. The square-topped audio waves generated this clipper are high in harmonic content, but these higher order harmonics may be greatby ly reduced by a low -level speech filter. Grid Limiters A triode grid limiter is shown in figure 3. On positive peaks of the input signal, the triode grid attempts to swing positive, and the grid- cathode resistance drops to a value on the order of 1000 ohms or so. The voltage drop across R (usually of the order of I megohm) is large compared to the grid-cathode drop, and the resulting limiting action removes the top part of the positive input wave. Clamping Circuits 10 -2 A circuit which holds either amplitude extreme of a waveform to a given reference level e OUT IN 187 eiN eouT DIODE CONDUCTS OA POSITIVE CLAMPING CIRCUIT SIMPLE POSITIVE © NEGATIVE CLAMPING CIRCUIT Figure 4 AND NEGATIVE CLAMPING CIRCUITS www.americanradiohistory.com 188 e Special Vacuum Tube Circuits r -, L- J RADIO THE ¡DEFLECTION COIL IN -100Y CI CHARGE PATH NEGATIVE PLOYED IN Figure 5 CLAMPING CIRCUIT EMELECTROMAGNETIC SWEEP C2 DISCHARGE PATH Figure 7 THE CHARGE AND DISCHARGE PATHS IN FREE -RUNNING MULTIVIBRATOR OF FIGURE 6 SYSTEM is repeated and therefore is "jittery." If a clamping circuit is placed between the sweep amplifier and the deflection element, the start of the sweep can be regulated by adjusting the d -c voltage applied to the clamping tube (fig- B+ ure 5). Multivibrators 10-3 Figure 6 BASIC MULTIVIBRATOR CIRCUIT The multivibrator, or relaxation oscillator, is used for the generation of nonsinusoidal waveforms. The output is rich in harmonics, but the inherent frequency stability is poor. The multivibrator may be stabilized by the introduction of synchronizing voltages of harmonic or subharmonic frequency. In its simplest form, the multivibrator is a simple two -stage resistance -capacitance coupled amplifier with the output of the second stage coupled through a capacitor to the grid of the first tube, as shown in figure 6. Since the output of the second stage is of the proper polarity to reinforce the input signal applied to the first tube, oscillations can readily take place, started by thermal agitation noise and of potential is called a clamping circuit or a d-c restorer. Clamping circuits are used after RC cpupling circuits where the wave f o r m swing is required to be either above or below the reference voltage, instead of alternating on both sides of it (figure 4). Clamping circuits are usually encountered in oscilloscope sweep circuits. If the sweep voltage does not always start from the same reference point, the trace on the screen does not begin at the same point on the screen each time the sweep B. B. NIP /// SYNCHNONIZING SIGNAL B DIRECT- COUPLED CATHODE MULTI VIBRATOR ELECTRON-COUPLED MULTIVIBRATOR Figure © MULTI VIBRATOR WITH SINE -WAVE SYNCHRONIZING SIGNAL APPLIED TO ONE TUBE 8 VARIOUS FORMS OF MULTIVIBRATOR CIRCUITS www.americanradiohistory.com - Multivibrators 189 PULSE OUTPUT ONE -SHOT MULTIVIBRATOR BASIC ECCLES-JORDAN TRIGGER CIRCUIT Figure 9 ECCLES -JORDAN MULTI VIBRATOR CIRCUITS ) miscellaneous tube noise. Oscillation is maintained by the process of building up and discharging the store of energy in the grid coupling capacitors of the two tubes. The charging and discharging paths are shown in figure 7. Various forms of multivibrators are shown in figure 8. The output of a multivibrator may be used as a source of square waves, as an electronic switch, or as a means of obtaining frequency division. Submultiple frequencies as low as one -tenth of the injected synchronizing frequency may easily be obtained. The Eccles- Jordan Circuit The Eccles -Jordan trigger circuit is shown in figure 9A. This is not a true mul- tivibrator, but rather a circuit that possesses two conditions of stable equilibrium. One condition is when V, is conducting and V2 is cutoff; the other when V2 is conducting and V, is cutoff. The circuit remains in one or the other of these two stable conditions with no change in operating potentials until some external action occurs which causes the nonconducting tube to conduct. The tubes then reverse their functions and remain in the new condition as long as no plate current flows in the cutoff tube. This type of circuit is known as a flip - flop circuit. Figure 9B illustrates a modified Eccles Jordan circuit which accomplishes a complete cycle when triggered with a positive pulse. Such a circuit is called a one -shot multivibrator. For initial action, V, is cutoff and V2 is conducting. A large positive pulse applied to the grid of V, causes this tube to conduct, and the voltage at its plate decreases by virtue of the IR drop through R3. Capacitor C2 is charged rapidly by this abrupt change in V, plate voltage, and V, becomes cutoff while V, conducts. This condition exists until C2 discharges, allowing V2 to conduct, raising the cathode bias of V, until it is once again cutoff. A direct, cathode -coupled multivibrator is shown in figure 8A. RK is a common cathode resistor for the two tubes, and coupling takes place across this resistor. It is impossible for a tube in this circuit to completely cutoff the other tube, and a circuit of this type is called a free- running multivibrator in which the condition of one tube temporarily cuts off the other. RF PULSE RF RF PULSE PULSE nns nnl '1 eoUT nnl CUTOFF TIME C ouT I_ Figure 10 BLOCKING TIME TIMG MNI3CIC04E CUTOFF TIME SINGLE -SWING CUTOFF OSCILLATOR Figure 11 HARTLEY OSCILLATOR USED AS BLOCKING OSCILLATOR BY PROPER CHOICE OF R, -C, 190 Special Vacuum eIN Tube Circuits e Pour POSITIVE COUNTING CIRCUIT THE eouT NEGATIVE COUNTING CIRC, POSITIVE '.ETEA Figure POSITIVE AND NEGATIVE 10 -4 The Blocking Oscillator A blocking oscillator is any oscillator which cuts itself off after one or more cycles caused by the accumulation of a negative charge on the grid capacitor. This negative charge may gradually be drained off through the grid resistor of the tube, allowing the circuit to oscillate once again. The process is repeated and the tube becomes an intermittent oscillator. The rate of such an occurance is determined by the R-C time constant of the grid circuit. A single -swing blocking oscillator is shown in figure 10, wherein the tube is cutoff before the completion of one cycle. The tube produces single pulses of energy, the time between the pulses being regulated by the discharge time of the grid R -C network. The self-pulsing blocking oscillator is shown in figure 11, and is used to produce pulses of r-f energy, the number of pulses being de- termined by the timing network in the grid circuit of the oscillator. The rate at which these pulses occur is known as the pulse -repetition frequency, or p.r. /. 10 -5 ADIO IN .. NG )N CIRCUIT WITH 12 COUNTING CIRCUITS ing units to be counted, and produces a- voltage that is proportional to the frequency of the pulses. A counting circuit may be used in conjunction with a blocking oscillator to produce a trigger pulse which is a submultiple of of the frequency of the applied pulse. Either positive or negative pulses may be counted. A positive counting circuit is shown in figure 12A, and a negative counting circuit is shown in figure 12B. The positive counter allows a certain amount of current to flow through R, each time a pulse is applied to C,. The positive pulse charges and makes the plate of V, positive with respect to its cathode. V, conducts until the exciting pulse passes. C, is then discharged by and the circuit is ready to accept another pulse. The average current flowing through R, increases as the pulse- repetition frequency increases, and decreases as the p.r.f. decreases. By reversing the diode connection s, as shown in figure 12B, the circuit is made to C V respond to negative pulses. In this circuit, an increase in the p.r.f. causes a decrease in the average current flowing through which is opposite to the effect in the positive counter. R Counting Circuits A counting circuit, or frequency divider is one which receives uniform pulses, represent- e Vs feIN 3 e OUTy Figure Figure 13 STEP -BY -STEP COUNTING CIRCUIT 14 The step -by -step counter used to trigger a blocking oscillator. The blocking oscillator serves as a frequency divider. www.americanradiohistory.com HANDBOOK R -C Oscillators 191 4 C2 CI !00'1. C VI R=LP RB LP R4. 4 a1 xCI =R2 THE WATT, 110 X V LAMP BULB Ca Figure 15 WIEN- BRIDGE AUDIO OSCILLATOR A step- counter is similar to the circuits discussed, except that a capacitor which is large compared to C, replaces the diode load resistor. The charge of this condenser is increased during the time of each pulse, producing a step voltage across the output (figure 13). A blocking oscillator may be connected to a step- counter, as shown in figure 14. The oscillator is triggered into operation when the voltage across C, reaches a point sufficiently positive to raise the grid of V, above cutoff. Circuit parameters may be chosen so that a count division up to 1/20 may be obtained with reliability. 10 -6 Resistance -Capacity Oscillators In an R -C oscillator, the frequency is determined by a resistance capacity network that provides regenerative coupling between the output and input of a feedback amplifier. No use is made of a tank circuit consisting of inductance and capacitance to control the frequency of oscillation. The Wien - Bridge oscillator employs a Wien network in the R -C feedback circuit and is shown in figure 15. Tube V, is the oscillator tube, and tube V, is an amplifier and phase inverter tube. Since the feedback voltage through C4 produced by V, is in phase with the input circuit of V, at all frequencies, oscillation is maintained by voltages of any frequency that exist in the circuit. The bridge circuit is used, then, to eliminate feedback voltages of all frequencies except the single frequency desired at the output of the oscillator. The bridge allows a voltage of only one frequency to be effective in the circuit because of the degeneration and phase shift provided by this Figure 16 PHASE -SHIFT OSCILLATOR THE circuit. The frequency at which oscillation occurs is: f- 1 , 2n R, C, when R,xC,=R,xC, Lp is used as the cathode resistor thermal stabilizer of the oscillator amplitude. The variation of the resistance with respect to current of the lamp bulb holds the oscillator output voltage at a nearly constant amplitude. The phase -shi /t oscillator shown in figure 16 is a single tube oscillator using a three section phase shift network. Each section of the network produces a phase shift in proportion to the frequency of the signal that passes through it. For oscillations to be produced, the signal from the plate of the tube must be shifted 180 °. Three successive phase shifts of 60° accomplish this, and the frequency of oscillation is determined by this phase shift A high -mu triode or a pentode must be used in this circuit. In order to increase the frequency of oscillation, either the resistance or the capacitance must be decreased. A lamp of V, as THE a Figure 17 BRIDGE -TYPE PHASE -SHIFT OSCILLATOR www.americanradiohistory.com 192 Special Vacuum Tube Circuits THE RADIO 0-FREQ. OF OSCILLATION NEC F/B =POS F/B -NOTCHFREQUENCY F- NEGATIVE I 2?RC FEEDBACK WHERE (LOOP C=1/Ct C2 2) POSITIVE FEEDBACK (LOOP r) f rFREQ OF OSCILLATION PHASE SHIFT'0 "NOTCH NETWORK Figure 19 BRIDGE -T FEEDBACK LOOP CIRCUITS Figure 18 THE NBS BRIDGE -T OSCILLATOR CIRCUIT AS USED IN THE HEATH AG -9 AUDIO GENERATOR A bridge -type phase shill oscillator is shown in figure 17. The bridge is so proportioned that at only one frequency is the phase shift through the bridge 180 °. Voltages of other frequencies are fed back to the grid of the tube out of phase with the existing grid signal, and are cancelled by being amplified out of phase. The NBS Bridge -T oscillator developed by the National Bureau of Standards consists of a two stage amplifier having two feedback loops, as shown in figure 18. Loop 1 consists of a regenerative cathode-to- cathode loop, consisting of Lp, and C3, The bulb regulates the positive feedback, and tends to stabilize the output of the oscillator, much as in the manner of the Wien circuit. Loop 2 consists of a grid -cathode degenerative circuit, containing the bridge -T. Oscillation will occur at the null frequency of the bridge, at which frequency the bridge allows minimum degeneration in loop 2 (figure 19). Oscillation will occur at the null frequency of the bridge, at which frequency the bridge allows minimum degeneration in loop 2. effect system. The furnace (F) raises the room temperature (T) to a predetermined value at which point the sensing thermostat (TAI) reduces the fuel flow to the furnace. When the room temperature drops below the predetermined value the fuel flow is increased by the and thermostat control. An interdependent control system is created by this arrangement: the room temperature depends upon the thermostat action, and the thermostat action depends upon the room temperature. This sequence of events may be termed a closed loop feedback system. ROOM FURNACE TEMPERATURE (F) (T) FEEDBACK (ERROR SIGNAL) 10 -7 Feedback discus Feedback amplifiers have been sed in Chapter 6, section 15 of this Handbook. A more general use of feedback is in automatic control and regulating systems. Mechanical feedback has been used for many years in such forms as engine speed governors and steering servo engines on ships. A simple feedback system for temperature control is shown in figure 20. This is a cause Figure 20 SIMPLE CLOSED LOOP FEEDBACK SYSTEM Room temperature (T) controls fuel supply to furnace (F) by feedloop through Thermostat back (TH) control. www.americanradiohistory.com Feedback HANDBOOK INPUT SIGNAL ¡OUTPUT SIGNAL PHASE SHIFT OF SYSTEM I! TIME OUTPUT SIGNAL FEEDBACK SIGNAL NO PHASE SHIFT .. . I A FEEDBACK SIGNAL WITH 180 PHASE SHIFT TIME Figure 21 PHASE SHIFT OF ERROR SIGNAL MAY CAUSE OSCILLATION INCLOSED LOOP SYSTEM To prevent oscillation, the gain of the feedback loop must be less than unity when the phase shift of the system reaches 180 degrees. Error Cancellation A feedback control system is dependent upon a degree of error in the output signal, since this error component is used to bring about the correction. This component is called the error signal. The error, or deviation from the desired signal 193 is passed through the feedback loop to cause an adjustment to reduce the value of the error signal. Care must be taken in the design of the feedback loop to reduce over -control tendencies wherein the correction signal would carry the sytem past the point of correct operation. Under certain circumstances the new error signal would cause the feedback control to overcorrect in the opposite direction, resulting in hunting or oscillation of the closed loop system about the correct operating point. Negative feedback control would tend to dampout spurious system oscillation if it were not for the time lag or phase shift in the system. If the overall phase shift is equal to one half cycle of the operating frequency of the system the feedback will maintain a steady state of oscillation when the circuit gain is sufficiently high, as shown in figure 21. In order to prevent oscillation, the gain figure of the feedback loop must be less than unity when the phase shift of the system reaches 180 degrees. In an ideal control system the gain of the loop would be constant throughout the operating range of the device, and would drop rapidly outside the range to reduce the bandwidth of the control system to a minimum. The time lag in a closed loop system may reduced by using electronic circuits in place of mechanical devices, or by the use of special circuit elements having a phase -lead characteristic. Such devices make use of the properties of a capacitor, wherein the current leads the voltage applied to it. be www.americanradiohistory.com CHAPTER ELEVEN Electronic Computers Mechanical computing machines were first produced in the seventeenth century in Europe although the simple Chinese abacus (a digital computer) had been in use for centuries. Until the last decade only simple mechanical computers (such as adding and bookkeeping machines) were in general use. The transformation and transmission of the volume of information required by modern CIIP 011, technology requires that machines assume many of the information processing systems formerly done by the human mind. Computing machines can perform routine operations more quickly and more accurately than a human being, processing mathematical and logistical data on a production line basis. The computer, however, cannot create, but can only follow instructions. If the instructions are in error, 0111 THE IBM COMPUTER AND "MEMORY" The puter "704" Comis used 32,000 with "word" storage memory unit for research Heart programs. of this auxiliary unit are small, doughnut- shaped iron ferrites which store information by means of magnetism. The unit is the first of Ms kind to be installed with I B M's 704 computer. Components of the system seen in the foreground are (left) card punch and (right) cord .j reader. In the center of the picture is the 704's processing unit. www.americanradiohistory.com Digital Computers 195 NORTN SNORE FARN!R CORN N!N ^FOX NOTE: ALL BUTTONS NAVE ONE NORMALLY OPEN CONTACT AND ONE NORMALLY CLOSED CONTACT. Figure 2 A SEQUENCE COMPUTER. Three correct buttons will sound the buzzer. SOUTN SNORE Figure 1 SIMPLE PUZZLES IN LOGIC MAY BE SOLVED BY ELECTRIC COMPUTER. THE "FARMER AND RIVER" COMPUTER IS SHOWN HERE. the computer will produce a wrong answer. Computers may be divided into two classes: the digital and the analog. The digital computer counts, and its accuracy is limited only by the number of significant figures provided for in the instrument. The analog computer measures, and its accuracy is limited by the percentage errors of the devices used, multiplied by the range of the variables they represent. 11 -1 Digital Computers The digital computer operates in discrete steps. In general, the mathematical operations are performed by combinations of additions. Thus multiplication is performed by repeated additions, and integration is performed by summation. The digital computer may be thought of as an "on -off" device operating from signals that either exist or do not exist. The common adding machine is a simple computer of this type. The "on -off" or "yes-no" type of situation is well suited to switches, electrical relays, or to electronic tubes. A simple electrical digital computer may be used to solve the old "farmer and river" problem. The farmer must transport a hen, a bushel of corn, and a fox across a river in a small boat capable of carrying the farmer plus one other article. If the farmer takes the fox in the boat with him, the hen will eat the corn. On the other hand, if he takes the corn, the fox will eat the hen. The circuit for a simple computer to solve this problem is shown in figure 1. When the switches are moved from "south shore" to "north shore" in the proper sequence the warning buzzer will not sound. An error of choice will sound the buzzer. A second simple "digital computer" is shown is figure 2. The problem is to find the three proper push buttons that will sound the buzzer. The nine buttons are mounted on a board so that the wiring cannot be seen. Each switch of these simple computers executes an "on -off" action. When applied to a logical problem "yes-no" may be substituted for this term. The computer thus can act out a logical concept concerned with a simple choice. An electronic switch (tube) may be substituted for the mechanical switch to increase the speed of the computer. The early computers, such as the ENIAC (Electronic Numerical Integrator and Calculator) employed over 18,000 tubes for memory and registering circuits capable of "remembering" a 10 -digit number. 11 -2 Binary Notation To simply and reduce the cost of the digital computer it was necessary to modify the system of operation so that fewer tubes were used per bit of information. The ENIAC -type computer requires 50 tubes to register a 5 -digit number. O 0 0 0 O O O O O O O O O O O O O O O O O O O {S. O O O O O O O O O O O O :;oFigure 0 O 0 0 O O O O 3 BINARY NOTATION MAY BE USED FOR DIGITAL DISPLAY. BINARY BOARD ABOVE INDICATES "73092." www.americanradiohistory.com THE RADIO Electronic Computers 196 O O -` :O"-- BINARY NOTATION 0 o TUBE(S) DIGIT I 1 a a 3 2+1 4 4 5 +1 7 +2+1 s 4 9 !+1 10 6+2 1/ 5+2+1 12 + +4+1 14 +4+2 15 +4+2+1 1 1 2 1,0 3 1,1 s 1,0.0 1,0,1 1.1.0 6 - 4+2 12 DECIMAL NOTATION , 7 1.1.1 0 1,0,0,0 1,0,0,1 1,0,1,0 0 10 Figure 5 BINARY NOTATION SYSTEM REQUIRES ONLY TWO NUMBERS, "0" AND "1." Figure 4 BINARY DECIMAL NOTATION. ONLY FOUR TUBES ARE REQUIRED TO REPRESENT DIGITS FROM TO 15. THE DIGIT "12" IS INDICATED 1 ABOVE. The tubes (or their indicator lamps) can be arranged in five columns of 10 tubes each. From right to left the columns represent units, tens, hundreds, thousands, etc. The bottom tube in each column represents "zero," the second tube represents "one," the third tube "two," and so on. Only one tube in each column is excited at any given instant. If the number 73092 is to be displayed, tube number seven in the fifth column is excited, tube number three in the fourth column, tube number zero in the third column, etc. as shown in figure 3. A simpler system employs the binary decimal notation, wherein any number from one to fifteen can be represented by four tubes. Each of the four tubes has a numerical value that is associated with its position in the tube group. More than one tube of the group may be excited at once, as illustrated in figure 4. The values assigned to the tubes in this particular group are 1, 2, 4, and 8. Additional tubes may be added to the group, doubling the notation of the rube thus: 1, 2, 4, 8, 16, 32, 64, 128, 356, etc. Any numerical value lower than the highest group number can be displayed by the correct tube combination. A third system employs the binary notation which makes use of a bit (binary digit) representing a single morsel of information. The binary system has been known for over forty centuries, and was considered a mystical revelation for ages since it employed only two sym- bols for all numbers. Computer service usually employs "zero" and "one" as these symbols. Decimal notation and binary notation for common numbers are shown in figure 5. The binary notation represents 4 -digit numbers (thousands) with .ten bits, and 7 -digit numbers (millions) with 20 bits. Only one electron tube is required to display an information bit. The savings in components and primary power drain of a binary -type computer over the older ENIAC -type computer is obvious. Figure 6 illustrates a computer board showing the binary indications from one to ten. Digital Computer The digital computer is em- ployed in a "yes -no" situation. It may be used for routine calculations that would ordinarily require enormous man -hours of time, such as checking stress estimates in aircraft design, or military logistics, and problems involving the manipulation of large masses of figures. Uses DECIMAL NOTATION COMPUTER NOTATION o O 1 o o a 3 0 5 0 o 4 O O o 0 0 0O O 4 7 . o 0 0 + 10 0 = Figure o OFF 6 BINARY NOTATION AS REPRESENTED ON COMPUTER BOARD FOR NUMBERS FROM 1 TO 10. www.americanradiohistory.com HANDBOOK Analog Computers eour=e,+e2 197 +150V. e Our R, e, eour- R, R, +R2 e2 R2 Ri +R2 / R3 R, R2 \[R,+ R234-R3J Figure 7 OF TWO VOLTAGES BY ELECTRICAL MEANS. SUMMATION Analog Computers 11 -3 The analog computer represents the use of one physical system as a model for a second system that is usually more difficult to construct or to measure, and that obeys the equations of the same form. The term analog implies similarity of relations or properties between the two systems. The common slide-rule is a mechanical analog computer. The speedometer in an automobile is a differential analog computer, displaying information proportional to the rate of change of speed of the vehicle. The electronic analog computer employs circuits containing resistance, capacitance, and inductance arranged to behave in accordance with analogous equations. Variables are represented by d -c voltages which may vary with time. Figure 8 OF TWO VOLTAGES BY ELECTRONIC MEANS. SUMMATION Thus complicated problems can be solved by d -c amplifiers and potentiometer controls in electronic circuits performing mathematical functions. If a linear network is energized by two voltage sources the voltages may be summed as shown in figure 7. Subtraction of quantities may be accomplished by using negative and positive voltages. A -c voltages may be employed for certain additive circuits, and more Addition and Subtraction THE HEATHKIT ELECTRONIC ANALOG DIGITAL COMPUTER "electronic slide rule" simuThis lates equations or physical problems electronically, sub- stituting one phys- ical system as a model for a sec- system that usually more or costly to construct or measure, and that obeys equations of the some form. ond is difficult www.americanradiohistory.com eour THE RADIO Electronic Computers 198 R2 . tt = c e eour Figure 9 ELECTRONIC MULTIPLICATION MAY BE ACCOMPLISHED BY CALIBRATED POTENTIOMETERS, WHEN OUTPUT VOLTAGE IS PROPORTIONAL TO THE INPUT VOLTAGE MULTIPLIED BY A CONSTANT (R R1). complex circuits employ vacuum tubes, as in figure 8. Synchronous transformers may be used to add expressions of angular rotation, and circuits have been developed for adding time delays, or pulse counting. Multiplication Electronic multiplication and division may be accomplished with the use of potentiometers where the output voltage is proportional to the input voltage multiplied by a constant which may be altered by changing the physical arrangement of the potentiometer (figure 9). Variable autotransformers may also be used to perform multiplication. A simple bridge may be used to obtain an output that is the product of two inputs divided by a third input, as shown in figure 10. and Division Differentiation The time derivative of a voltage can be expressed as a charge on a capacitor by: de (1) dt and is shown in figure 11A. The charging current is converted into a voltage by the use of a resistor, R. If the input to the RC circuit is charging at a uniform rate so that the current through C and R is constant, the output voltage e is: i OUTPUT'\, =C e our Figure 11 ELECTRONIC DIFFERENTIATION The time derivative of a voltage can be expressed as a charge on a capacitor (A). Operational amplifier (B) employs feed back principle for short differentiation time. RCd (2) For highest accuracy, a small RC product should be used, permitting the maximum possible differentiation time. The output of the differentiator may be amplified to any suitable level. A more accurate differentiating device makes use of an operational amplifier. This unit is a high gain, negative feedback d -c amplifier (discussed in section 11 -4) with the resistance portion of the RC product appearing in the feedback loop of the amplifier (figure 11B). A shorter differentiation time may be employed if the junction point between R and C could be held at a constant potential. The feedback amplifier shown inverts the output signal and applies it to the RC network, hold- ing the junction potential constant. Integration Integration is a process of accumulation, or summation, and requires a device capable of storing physical quantities. A capacitor will store an electrical charge and will give the time integral of a current in respect to a voltage: eo= 1 idt (3) In most computers, the input signal is in the form of a voltage, and the input charging current of the capacitor must be taken through a series resistance as in figure 12. If the integrating time is short the charging current is approximately proportional to the input voltage. The charging current may be made a true measure of the input voltage by the use RI x R3 R2 Figure 10 ELECTRONIC MULTIPLICATION BY BRIDGE CIRCUIT PROVIDES OUTPUT THAT IS PRODUCT OF TWO INPUTS DIVIDED BY A THIRD INPUT. Figure 12 SIMPLE INTEGRATION CIRCUIT Making use of charging current of capacitor. www.americanradiohistory.com e O T c eouT HANDBOOK Operational Amplifier - The Operational -4 11 199 Amplifier Mathematical operations are performed by using a high gain d -c amplifier, termed an operational amplifier. The symbol of this unit is a triangle, with the apex pointing in the direction of operation ( figure 15). The gain of such an amplifier is -A, so: + Figure 3 "MILLER FEEDBACK" INTEGRATOR SUITABLE FOR COMPUTER USE. eaur eo=-Aeg,oreg= Figure 14 R -L NETWORK USED FOR INTEGRATION PURPOSES. -e (4) A If -A approaches infinity, e, will be approximately zero. In practice this condition is realized by using amplifiers having open loop gains of 30,000 to 60,000. If ea is set at 100 volts, e, will be of the order of a few millivolts. Thus, considering eg equal to zero: o Figure 15 OPERATIONAL AMPLIFIER ( -A) Mathematical operations may be performed by any operational amplifier, usually a stable, high-gain d -c amplifier, such as RI - Rf ,oreo- Rr RI (5) eI which may be written: shown in Figure 16. e0 of an operational amplifier wherein the capacitance portion of the RC product appears in the feedback loop of the amplifier, holding the junction point between R and C at a constant potential. A simple integrator is shown in figure 13 employing the Miller feedback principle. Integration is also possible with an RL network (figure 14). _ - Keg, where K = R I (6 ) This amounts to multiplication by a constant coefficient, since RI and Rr may be fixed in value. The circuit of a typical operational amplifier is shown in figure 16. Amplifier Operation Two voltages may be added by the amplifier, as shown in figure 17. Keeping in mind that eg is u (Rv) e, BIAS -4í0V GAIN= -ze0 V -A Figure 16 HIGH GAIN OPERATIONAL AMPLIFIER, SUCH AS USED IN HEATH COMPUTER. 200 THE RADIO Electronic Computers RF o e ea D o eo Figure 17 TWO VOLTAGES MAY BE ADDED BY SUMMATION AMPLIFIER. F essentially at zero (ground) potential: eo= eo or, where Re = R. K. K= e Re + e,+K:e (8) and K: = B (7) e2 R3 R` As long as e., does not exceed the input range of the amplifier, any number of inputs may be used: eo= - Re Rl ei Rf ei + + Á= - - - + RI Ro en 11 By combining the above operations in various ways, problems of many kinds may be solved .For example, consider the mass- springdamper assembly shown in figure 19. The mass M is connected to the spring which has an elastic constant K. The viscous damping constant is C. The vertical displacement is y. The sum of. the forces acting on mass M is: f or eo = -Rr+ (11) Ro Integration is performed by replacing the feedback resistor Re with a capacitor Ce, as shown in figure 18. For this circuit (with ea approximately zero): t but g=Cr en, so dc ei dt Ri d= Ce do (12) and R = Cf 1 and o e o R 1Cr eidt deo eo ei dt Ri Cr (t) =M d$ + C d +Ky (14) (15) . M dt' -C dt -Ky +f (t) (17) If, in the analog circuit, there is a voltage -dt equal to M it can be converted to dby passing it through an integrator circuit having an RC time constant equal to M. This resulting voltage can be passed through a second integrator stage with unit time constant which will haue an output volage equal to y. The voltages representing y, Figure 18 INTEGRATION RI AN oeo o Performed by Summation Amplifier by replacing feed back resistor with a capacitor. (16) where f (t) is the applied force, or forcing function. The first step is to set up the analog computer circuit so as to obtain an output voltage proportional to y for a given input voltage proportional to f (t) Equation (16) may be rewritten in the form: deo (13) Thus: Solving Analog Problems -5 (10) en =eft/ Figure 19 "MASS- SPRINGDAMPER" PROBLEM MAY BE SOLVED BY ELECTRICAL ANALOGY WITH SIMPLE COMPUTER. then be summed to give (t) can - Ky + f (t) and f -C do which is the right hand side of equation (17) , r and therefore equal to M www.americanradiohistory.com dt . Connecting the HANDBOOK Analog Problems 201 SET VOLTAGE TO Y =YO As TIME (INITIAL t'0 M dZ -2.dr -Cdt -Kr+f(t dt dt 1 DISPLACEMENT) DISPLAY OSCILLOSCOPE TO Cdt - fit) F(t) Figure 20 ANALOG SOLUTION FOR "MASS- SPRINGDAMPER" PROBLEM OF FIGURE 19. output of the summing amplifier (A3) to the input of the first as shown in figure 20 satisfies the equation. To obtain a solution to the problem, the initial displacement and velocity must be specified. This is done by charging the integrating capacitors to the proper voltages. Three operational amplifiers and a summing amplifier are required. A second problem that may be solved by the analog computer is the example of a freely falling body. Disregarding air resistance, the body will fall (due to the action of gravity) with a constant acceleration. The equation describing this action is: F =mg =m (18) dt' Integration of equation (18) will give the velocity, or , dt and integration a second time will give displacement, or y. The block diagram of a suitable computer for this problem is shown in figure 21. If a voltage proportional to g and hence to d' dY is introduced into the first amplifier, the - output of that unit will he -H , or the --R2 Ri e,= dt dZr dtZ r -dr dt eo G Figure 21 ANALOG COMPUTER FOR "FREELY FALLING BODY" PROBLEM. velocity. That, in turn, will become y, or distance, at the output of the second amplifier. Before the problem can be solved on the computer it is necessary to determine the time of solution desirable and the output amplitude of the solution. The time of solution is determined by the RC constant of the integrating amplifiers. If RC is set at unity, computer time is equal to real time. The computer time desirable is determined by the method of readout. When using an oscilloscope for read -out, a short solution time is desirable. For a recorder, longer solution time is better. Suppose, for example, in the problem of the falling body, the distance of fall in 2.5 seconds is desired. Using an RC constant of 1 would give a solution time of 2.5 seconds. This would be acceptable for a recorder but is slow for an oscilloscope. A convenient time of solution for the 'scope would be 25 milliseconds. This is 1/100 of the real time, so an RC constant of .01 is needed. This can be obtained with C equal to 0.1 pfd, and R equal to 100,000 ohms. It is now necessary to choose an input voltage which will not overdrive the amplifiers. The value of g is known to be approximately 32 ft /sec. /sec. A check indicates that if we set g equal to 32 volts, the voltage representing the answer will exceed 100 volts. Since the linear response of the amplifier is only 100 volts, this is undesirable. An input of 16 volts, however, should permit satisfactory operation of the amplifiers. Output voltages near zero should also be avoided. In general, output voltage should be about 50 volts or so, with amplifier gains of 20 to 60 being preferable. Thus, for this particular problem the time scale factor and amplitude -scale factor have www.americanradiohistory.com 202 THE RADIO Electronic Computers A' A RELAY RELAY 100 K d2r 16 V. dt2 dr eo= Y Figure 22 ANALOG SOLUTION FOR "FALLING BODY PROBLEM" OF FIGURE 21. Figure 24 LIMITING CIRCUIT TO SIMULATE NON -LINEAR FUNCTIONS SUCH AS TIME IN SECONDS 4 eo e, tr dt (c ENCOUNTERED IN HYSTERESIS, BACKLASH, AND FRICTION PROBLEMS. s) 6 16 eo 32 VOLTS DISTANCE N 11 -6 FEET -24 46 1 -32 Problems are frequently encountered in which non -linear functions must be simulated. Non -linear potentiometers may be used to supply an unusual voltage source, or diodes may be used as limiters in those problems in which a function is defined differently for different regions of the independent variable. Such a function might be defined as follows: 64 -40 -80 ) _'98 -46 36 1'00) 112 64 28 72 ,44 Non -linear Functions Figure 23 READ -OUT SOLUTION OF "FREELY FALLING BODY" PROBLEM. e.. e.. e.. been chosen. The problem now looks like figure 22. To solve the problem, relays A and A' are opened. The solution should now appear on the oscilloscope as shown in figure 23. The solution of the problem leaves the integrating capacitors charged. It is necessary to remove this charge before the problem can be rerun. This is done by closing relays A and A'. -- - K: = K1, e, = et, KT,, e_, = K2, el ,, K: KI (19) (20) (21) where K, and K: are constants. Various limiting circuits can be used, one of which is shown in figure 24. This is a series limiter circuit which is simple and does not require special components. Commonly encountered problems requiring these or similar limiting techniques include hysteresis, backlash, and certain types of friction. C NOTE: REPLACE C WITN A I MEC. RESISTOR FOR FUNCTION SETUP MEG X OUTPUT Y OUTPUT RAMP - FUNCTION GENERATOR 620 K P SLOPE CONTROL E 1 2 6AL5 BREAK CONTROL VOLTAGE 620K t SIGN CHANGING AMPLIFIER SUMMING AMPLIFIER Figure 25 SIMPLIFIED DIAGRAM OF FUNCTIONAL GENERATOR TO APPROXIMATE NON- LINEAR FUNCTIONS. HANDBOOK XI o Xz Non -Linear Functions 7X3 Figure 26 TYPICAL NON -LINEAR FUNCTION WHICH MAY BE SET UP WITH FUNCTION GENERATOR. The Function Generator function generator may be used to approximate almost any non -linear function. This is done by use of straight line segments which are combined to approximate curves such as are found in trigonometric functions as well as in stepped functions. In a typical generator ten line segments are used, five in the plus -x direction, and five in the minus -x direction. Five 6ÁL5 double diodes are used. Each line segment is generated by a modified bridge circuit (figure 25). A ramp function or voltage is fed into one arm of the bridge while the opposite arm is connected to a biased diode. The other two arms of the bridge combine to form the output. The voltage appearing across one of these arms is fed through a sign- changing amplifier and then summed with the voltage appearing at the opposite arm. If the arm of potentiometer P (the slope control) is set in the center, the bridge will be balanced and A Figure 27 ELECTRONIC PACKAGE IN DIGITAL COMPUTER. Stylized diagram of tube package. Lines carrying negative pulses are marked by a small circle at each end. Gates are indicated by a semi -circle with "pins" for each input. the output of the summing amplifier will be zero. If, on the other hand, potentiometer P is adjusted one way or the other from center, the bridge will be unbalanced and the summing amplifier output will vary linearly with respect to the input in either a positive or negative y direction, depending upon which side of center potentiometer P is set. The break voltage, or value of x at which a straight line segment will begin is set by biasing the diode to the particular voltage level or value of x desired. The ramp function generator has either a positive or negative input which because of the 180 degree phase shift in the amplifier, gives a minus- or plus -x output respectively. A typical function such as shown in figure 26 may be set up with the function generator. The initial condition volt- IBM's new "608," the first completely transistorized calculator for commercial applications, operates without the use of a single vacuum tube. Transistors - -tiny germanium devices that perform many of the functions of conventional vacuum tubes -make possible 50% reduction in computer -unit size and a in reduction 90% requirements power over a comparable IBM tube -model machine. They 203 are mounted, along with related circuitry, on banks of printed wiring panels in the 608. The machine's interstorage, or "memnal ory," is made up of magnetic cores -minute, doughnut -shaped objects that can "remember" information indefinitely, and recall it for use in calculations in a few millionths of a second. www.americanradiohistory.com 204 THE RADIO Electronic Computers age is set to the value of X. The break -voltage control is increased until the output of the summing amplifier increases abruptly, indicating the diode is conducting. The input voltage from the initial condition power supply is set to the value X: The slope control (P) is now set to value Y2. A second function generator may be used to set points X.1 and X., using the break -voltage control and the initial condition voltage adjustments. Points XS and X. are finally set with a third generator. The x- output of the function generator system may be read on an oscilloscope, using the x- output of the ramp- function generator amplifier as the horizontal sweep for the oscilloscope. Digital Circuitry 11 -7 Digital circuits dealing with "and," "or," and "not" situations may be excited by electrical pulses representing these logical operations. Sorting and amplifying the pulses can be accomplished by the use of electronic packages, such as shown in figure 27. Logical operations may be accomplished by diode -resistor gates operating into an amplifier stage. Negative and positive output pulses from the amplifier are obtained through diode output gates. The driving pulses may be obtained from a standard oscillator, operating at or near 1 mc. A circuit of a single digital package is shown in figure 28. Other configurations, such as a "flip-flop" may be used. Many such packages "AND can be connected in series to form operational circuits. The input "and" and "or" gates are biased to conduction by external voltages. The "and" diode gate transmits a pulse only when all the input terminals are pulsed positively, and the "or" diode gate transmits a positive pulse applied to any one of its input terminals. The input pulses pass through the gates and drive the amplifier stage, which delivers an amplified pulse to the positive and negative output gates, and to accompanying memory circuits. Memory Circuits A memory circuit consists of some sort of delay line which is capable of holding an information pulse for a period of time. The amount of delay is proportional to the frequency of the input signal. A "long" transmission line may be used as a delay line with the signal being removed from the "far" end of the line after being delayed an interval equal to the time of transmission along the line. Lines of this type are constructed in the manner of a coaxial cable, except that the inner conductor is a long, thin coil of wire. Other memory circuits make use of magnetostrictive or piezoelectric effects to retard the pulse. Information may also be stored in electrostatic storage tubes, upon magnetic recording tape, and in ferro- magnetic cores capable of holding 10,000 bits of information. GATES INPUT SI LIMITING INPUT DIODES 2 "OR" GATE INPUTSS INPUT *4 INPUT *5 INPUTS 6 INPUT* 7 CLAMPING DIODES 6AN5 POSITIVE PULSE 0 NEGATIVE PULSE INPUT Figure 28 TYPICAL DIGITAL PACKAGE SHOWING INPUT AND OUTPUT DIODE GATES AND PULSE AMPLIFIER. www.americanradiohistory.com CHAPTER TWELVE Radio Receiver Fundamentals A conventional reproducing device such as loudspeaker or a pair of earphones is incapable of receiving directly the intelligence carried by the carrier wave of a radio transmitting station. It is necessary that an additional device, called a radio receiver, be placed between the receiving antenna and the loudspeaker or headphones. Radio receivers vary widely in their complexity and basic design, depending upon the intended application and upon economic factors. A simple radio receiver for reception of radiotelephone signals can consist of an earphone, a silicon or germanium crystal as a carrier rectifier or demodulator, and a length of wire as an antenna. However, such a receiver is highly insensitive, and offers no significant discrimination between two signals in the same portion of the spectrum. On the other hand, a dual -diversity receiver designed for single -sideband reception and employing double or triple detection might occupy several relay racks and would cost many thousands of dollars. Ilowever, conventional communications receivers are intermediate in complexity and performance between the two extremes. This chapter is devoted to the principles underlying the operation of such conventional communications receivers. 12 -1 a Detection or Demodulation A detector or demodulator is a device for removing the modulation (demodulating) or detecting the intelligence carried by an incoming radio wave. Figure 1 illustrates an elementary form of radiotelephony receiver employing a diode detector. Energy from a passing radio wave will induce a voltage in the antenna and cause a radio- frequency current to flow from antenna to ground through coil Lt. The alternating magnetic field set up around L, links with the turns of L2 and causes an r -f current to flow through the parallel -tuned circuit, 1..2-C1. %hen variable capacitor C, is adjusted so that the tuned circuit is resonant at the frequency of the applied signal, the r-f voltage is maximum. This r -f voltage is applied to the diode detector where it is rectified into a varying direct current and passed through the earphones. The variations in this current correspond to the voice modulation placed on the signal at the transmitter. As the earphone diaphragms vibrate back and forth in accordRadiotelephony Demodulation 205 www.americanradiohistory.com 206 Radio Receiver Fundamentals THE RADIO TRIODE lquirlli 'Ili O AUDIO OUTPUT - GROUND L+ L2 I- e + PLATE- TICKLER REGENERATION WITH "THROTTLE' CONDENSER REGENERATION CONTROL. Figure ELEMENTARY FORM OF RECEIVER This is the basis of the "crystal set" type of 1 PENTODE ceiver, although a vacuum diode may be used in place of the crystal diode. The tank circuit L2 -C1 is tuned to the frequency it is desired to receive. The bypass capacitor across the phones should have a low reactance to the carrier frequency being received, but a high reactance to the modulation on the received signal. ance with the pulsating current they audibly reproduce the modulation which was placed upon the carrier wave. The operation of the detector circuit is shown graphically above the detector circuit in figure 1. The modulated carrier is shown at A, as it is applied to the antenna. B represents the same carrier, increased in amplitude, as it appears across the tuned circuit. In C the varying d -c output from the detector is seen. Radiotelegraphy Reception AUDIO OUTPUT re- +e -e CATHODE-TAP REGENERATION WITH SCREEN VOLTAGE REGENERATION CONTROL. Figure 2 REGENERATIVE DETECTOR CIRCUITS Regenerative detectors are seldom used at the present time due to their poor selectivity. However, they do illustrate the simplest type of receiver which may be used either for radiophone or radiotelegraph reception. Since a c -w telegraphy sig- nal consists of an unmodulated carrier which is interrupted to form dots and dashes, it is apparent that such a signal would not be made audible by detection alone. While the keying is a form of modulation, it is composed of such low frequency components that the keying envelope itself is below the audible range for hand keying speeds. Some means must be provided whereby an audible tone is heard while the unmodulated carrier is being received, the tone stopping immediately when the carrier is in- terrupted. The most simple means of accomplishing this is to feed a locally generated carrier of a slightly different frequency into the same detector, so that the incoming signal will mix with it to form an audible beat note. The difference frequency, or heterodyne as the beat note is known, will of course stop and start in accordance with the incoming c -w radiotelegraph signal, because the audible heterodyne can exist only when both the incoming and the locally generated carriers are present. The Autodyne Detector The local signal which is used to beat with the desired c -w signal in the detector may be supplied by a separate low-power oscillator in the receiver itself, or the detector may be made to self -oscillate, and thus serve the dual purpose of detector and oscillator. A detector which self -oscillates to provide a beat note is known as an autodyne detector, and the process of obtaining feedback between the detector plate and grid is called regeneration. An autodyne detector is most sensitive when it is barely oscillating, and for this reason a regeneration control is always included in the circuit to adjust the feedback to the proper amount. The regeneration control may be either a variable capacitor or a variable resistor, as shown in figure 2. With the detector regenerative but not oscillating, it is also quite sensitive. When the circuit is adjusted to operate in this manner, modulated signals may be received with considerably greater strength than with a non regenerative detector. www.americanradiohistory.com Superregenerative Detectors HANDBOOK 12 -2 Superregenerative Receivers ultra -high frequencies, when it is desired to keep weight and cost at a minimum, a special form of the regenerative receiver known as the superregenerator is often used for radiotelephony reception. The superregenerator is essentially a regenerative receiver with a means provided to throw the detector rapidly in and out of oscillation. The frequency at which the detector is made to go in and out of oscillation varies with the frequency to be received, but is usually between 20,000 and 500,000 times a second. This superregenerative action considerably increases the sensitivity of the oscillating detector so that the usual "background hiss" is greatly amplified when no signal is being received. This hiss diminishes in proportion to the strength of the received signal, loud signals eliminating the hiss entirely. TO At There are two systems in common use for causing the detector to break in and out of oscillation rapidly. In one, a separate interruption -frequency oscillator is arranged so as to vary the voltage rapidly on one of the detector tube elements (usually the plate, sometimes the screen) at the high rate necessary. The interruption- frequency oscillator commonly uses a conventional tickler- feedback circuit with coils appropriate for its operating frequency. The second, and simplest, type of super regenerative detector circuit is arranged so as to produce its own interruption frequency oscillation, without the aid of a separate tube. The detector tube damps (or "quenches ") itself out of signal- frequency oscillation at a high rate by virtue of the use of a high value of grid leak and proper size plate- blocking and grid capacitors, in conjunction with an excess of feedback. In this type of "self- quenched" detector, the grid leak is quite often returned to the positive side of the power supply (through the coil) rather than to the cathode. A representative self-quenched superregenerative detector circuit is shown in figure 3. Except where it is impossible to secure sufficient regenerative feedback to permit superregeneration, the self-quenching circuit is to be preferred; it is simpler, is self- adjusting as regards quenching amplitude, and can have good quenching wave form. To obtain as good results with a separately quenched superregenerator, very careful design is required. However, separately quenched circuits are useful when it is possible to make a certain tube oscillate on a very high frequency but it is impossible to obtain enough regeneration for self-quenching action. Quench Methods 2J7 AUDIO AMPLIFIER Figure 3 SUPERREGENERATIVE DETECTOR CIRCUIT A sell -quenched superregenerative detector such as illustrated above is capable of giving good sensitivity in the v-h -f range. However, the circuit has the disadvantage that its selectivity is relatively poor. Also, such o circuit should be preceded by an r -I stage to suppress the radiation of o signal by the oscillating detector. The optimum quenching frequency is a function of the signal frequency. As the operating frequency goes up, so does the optimum quenching frequency. hen the quench frequency is too low, maximum sensitivity is not obtained. When it is too high, both sensitivity and selectivity suffer. In fact, the optimum quench frequency for an operating frequency below 15 Mc. is in the audible range. This makes the superregenerator impracticable for use on the lower frequencies. The high background noise or hiss which is heard on a properly designed superregenerator when no signal is being received is not the quench frequency component; it is tube and tuned circuit fluctuation noise, indicating that the receiver is extremely sensitive. A moderately strong signal will cause the background noise to disappear completely, because the superregenerator has an inherent and instantaneous automatic volume control characteristic. This same a -v-c characteristic makes the receiver comparatively insensitive to impulse noise such as ignition pulses -a highly desirable feature. This characteristic also results in appreciable distortion of a received radiotelephone signal, but not enough to affect the intelligibility. The selectivity of a superregenerator is rather poor as compared to a superheterodyne, but is surprisingly good for so simple a receiver when figured on a percentage basis rather than absolute kc. bandwidth. FM Reception A. superregenerative receiver will receive frequency modulated signals with results comparing favorably with amplitude modulation if the frequency swing of the FM transmitter is sufficiently high. For such reception, the receiver is detuned slightly to either side of resonance. www.americanradiohistory.com 208 Radio Receiver Fundamentals THE RADI O AUDIO OUTPUT 22 MC OUTPUT IINTERMED. RF AMPLIFIER I "SECOND* REOUENCV AMPLIFIER I_ AUDIO 'AMPLIFIER DETECTOR 1 I - 0+100V FREONCY IOSCILLATORI (FOR C.W.) I AUOIO Figure 5 ESSENTIAL UNITS OF A SUPERHETERODYNE RECEIVER The basic portions of the receiver are shown in solid blocks. Practicable receivers employ the dotted blocks and also usually include such additional circuits os a noise limiter, Figure 4 THE FREMODYNE SUPERREGENERATIVE SUPERHETERODYNE DETECTOR FOR FREQUENCY MODULATED SIGNALS Superregenerative receivers radiate a strong, broad, and rough signal. For this reason, it is necessary in most applications to employ a radio frequency amplifier stage ahead of the detector, with thorough shielding throughout the receiver. The Fremodyne Detector The Hazel tin e- Fremodyne superregenerative circuit is expressly designed for re- ception of FM signals. This versatile circuit combines the action of the superregenerative receiver with the superhetrodyne, converting FM signals directly into audio signals in one double triode tube (figure 4). One section of the triode serves as a superregenerative mixer, producing an i -f of 22 Mc., an i -f amplifier, and a FM detector. The detector action is accomplished by slope detection tuning on the side of the i -f selectivity curve. This circuit greatly reduces the radiated signal, characteristic of the superregenerative detector, yet provides many of the desirable features of the superregenerator. The pass band of the Fremodyne detector is about 400 kc. 12 -3 Superheterodyne Receivers Because of its superiority and nearly universal use in all fields of radio reception, the circuit, and a crystal in the i -f amplifier. on a -v -c filter theory of operation of the superheterodyne should be familiar to every radio student and experimenter. The following discussion concerns superheterodynes for amplitude- modulation reception. It is, however, applicable in part to receivers for frequency modulation. Principle of In the superheterodyne, the incoming signal is applied to a mixer consisting of a non -linear impedance such as a vacuum tube or a diode. The signal is mixed with a steady signal generated locally in an oscillator stage, with the result that a signal bearing all the modulation applied to the original signal but of a frequency equal to the difference between the local oscillator and incoming signal frequencies appears in the mixer output circuit. The output from the mixer stage is fed into a fixed tuned intermediate -frequency amplifier, where it is amplified and detected in the usual manner, and passed on to the audio amplifier. Figure 5 shows a block diagram of the fundamental superheterodyne arrangement. The basic components are shown in heavy lines, the simplest superheterodyne consisting simply of these three units. However, a good communications receiver will comprise all of the elements shown, both heavy and dotted blocks. Operation Superheterodyne Advantages The advantages of super heterodyne reception are directly attributable to the use of the fixed -tuned intermediate -frequency (i -f) amplifier. Since all signals are converted to the intermediate frequency, this section of the receiver may be designed for optimum selectivity and high amplification. High amplification is easily obtained in the intermediatefrequency amplifier, since it operates at a www.americanradiohistory.com HANDBOOK The advantage over the tuned radio frequency (t -r -f) type of receiver because of what is commonly known as arithmetical selectivity. This can best be illustrated by considering two receivers, one of the t -r -f type and one of the superheterodyne type, both attempting to PENTODE 1 To BY -PASS MEG. CAPACITORS 05 TO 0.1 JIPD. A V C Figure 6 TYPICAL I -F AMPLIFIER STAGE relatively low frequency, where conventional pentode-type tubes give adequate voltage gain. A typical i -f amplifier is shown in figure 6. From the diagram it may be seen that both the grid and plate circuits are tuned. The tuned circuits used for coupling between i -f stages are known as i-I transformers. These will be more fully discussed later in this chapter. Choice of Intermediate Frequency The choice of a frequency for the i -f amplifier involves several considera- tions. One of these considerations concerns selectivity; the lower the intermediate frequency the greater the obtainable selectivity. On the other hand, a rather high intermediate frequency is desirable from the standpoint of image elimination, and also for the reception of signals from television and FM transmitters and modulated self -controlled oscillators, all of which occupy a rather wide band of frequencies, making a broad selectivity characteristic desirable. Images are a pecularity common to all superheterodyne receivers, and for this reason they are given a detailed discussion later in this chapter. While intermediate frequencies as low as 50 kc. are used where extreme selectivity is a requirement, and frequencies of 60 Mc. and above are used in some specialized forms of receivers, most present -day communications superheterodynes use intermediate frequencies around either 455 kc. or 1600 kc. Home -type broadcast receivers almost always use an i -f in the vicinity of 455 kc., while auto receivers usually use a frequency of about 262 kc. The standard frequency for the i -f channel of FM receivers is 10.7 Mc. Television receivers use an i -f which covers the band between about 21.5 and 27 Mc., although a new band between 41 and 46 Mc. is coming into more common usage. Arithmetical Selectivity 209 an overwhelming VARIABLE-1/ INPIp Superhetrodyne Aside from allowing the use of fixed -tuned band -pass amplifier stages, the superheterodyne has receive a desired signal at 10,000 kc. and eliminate a strong interfering signal at 10,010 kc. In the t -r -f receiver, separating these two signals in the tuning circuits is practically impossible, since they differ in frequency by only 0.1 per cent. However, in a superheterodyne with an intermediate frequency of, for example, 1000 kc., the desired signal will be converted to a frequency of 1000 kc. and the interfering signal will be converted to a frequency of 1010 kc., both signals appearing at the input of the i -f amplifier. In this case, the two signals may be separated much more readily, since they differ by 1 per cent, or 10 times as much as in the first case. The converter stage, or mixer, of a superheterodyne receiver can be either one of two types: (1) it may use a single envelope converter tube, such as a 6K8, 6SA7, or 6BE6, or (2) it may use two tubes, or two sets of elements in the same envelope, in an oscillator -mixer arrangement. Figure 7 shows a group of circuits of both types to illustrate present practice with regard to types of converter stages. Converter tube combinations such as shown in figures 7A and 7B are relatively simple and inexpensive, and they do an adequate job for most applications. With a converter tube such as the 6SB7 -Y or the 6BA7 quite satisfactory performance may be obtained for the reception of relatively strong signals (as for example FM broadcast reception) up to frequencies in excess of 100 Mc. However, the equivalent input noise resistance of such tubes is of the order of 200,000 ohms, which is a rather high value indeed. So such tubes are not suited for operation without an r -f stage in the high frequency range if weak -signal reception is The Converter Stage desired. The 6L7 mixer circuit shown in figure 7C, and the 6BA7 circuit of figure 71), also are characterized by an equivalent input noise re- sistance of several hundred thousand ohms, so that these also must be preceded by one or more r-f stages with a fairly high gain per stage if a low noise factor is desired of the complete receiver. However, the circuit arrangements shown at figures 7F and 6F are capable of low -noise operation within themselves, so that these circuits may be fed directly from the antenna without an r -f stage and still provide a good noise factor to the complete receiver. Note www.americanradiohistory.com 210 Radio Receiver THE Fundamentals RADIO 6SÁ7, 6SB7Y, roar AMP 6BE6. 6BÁ7 +250 2 V ULF + 50 V. Figure 7 TYPICAL FREQUENCY- CONVERTER (MIXER) STAGES The relative advantages of the different circuits are discussed in the text that both these circuits use control -grid injection of both the incoming signal and the local- oscillator voltage. Hence, paradoxically, circuits such as these should be preceded by an r -f stage if local- oscillator radiation is to be held to any reasonable value of field intensity. As the frequency of operation of a superheterodyne receiver is increased above a few hundred megacycles the signal -to -noise ratio appearing in the plate circuit of the mixer tube when triodes or pentodes are employed drops to a prohibitively low value. At frequencies above the upper -fre- quency limit for conventional mixer stages, mixers of the diode type are most commonly employed. The diode may be either a vacuum tube heater diode of a special u -h -f design such as the 9005, or it may be a crystal diode of the general type of the 1N21 through 1N28 series. Diode Mixers 12 -4 Mixer Noise' and Images The effects of mixer noise and images are troubles common to all superheterodynes. Since www.americanradiohistory.com HANDBOOK Mixer Characteristics both these effects can largely be obviated by the same remedy, they will be considered together. Mixer Noise Mixer noise of the shot- effect type, which is evidenced by a hiss in the audio output of the receiver, is caused by small irregularities in the plate current in the mixer stage and will mask weak signals. Noise of an identical nature is generated in an amplifier stage, but due to the fact that the conductance in the mixer stage is considerably lower than in an amplifier stage using the same tube, the proportion of inherent noise present in a mixer usually is considerably greater than in an amplifier stage using a comparable tube. Although this noise cannot be eliminated, its effects can be greatly minimized by placing sufficient signal- frequency amplification having a high signal -to -noise ratio ahead of the mixer. This remedy causes the signal output from the mixer to be large in proportion to the noise generated in the mixer stage. Increasing the gain after the mixer will be of no advantage in eliminating mixer noise difficulties; greater selectivity after the mixer will help to a certain extent, but cannot be carried too far, since this type of selectivity decreases the i -f band -pass and if carried too far will not pass the sidebands that are an essential part of a voice -modulated signal. A triode having a high trans conductance is the quietest mixer tube, exhibiting somewhat less gain but a better signal -to -noise ratio than a comparable multi -grid mixer tube. However, below 30 Mc. it is possible to construct a receiver that will get down to the atmospheric noise level without resorting to a triode mixer. The additional difficulties experienced in avoiding pulling, undesirable feedback, etc., when using a triode with control -grid injection tend to make multi -grid tubes the popular choice for this application on the lower frequencies. On very high frequencies, where set noise rather than atmospheric noise limits the weak signal response, triode mixers are more widely used. A 6J6 miniature twin triode with grids in push -pull and plates in parallel makes an excellent mixer up to about 600 Mc. Triode Mixers The amplitude of the injection volt age will affect the conversion trans conductance of the mixer, and therefore should be made optimum if maximum signal -to -noise ratio is desired. If fixed bias is employed on the injection grid, the optimum injection voltage is quite critical. If cathode bias is used, the optimum voltage is not so critical; and if grid leak bias is employed, the Injection Voltage 211 optimum injection voltage is not at all critical just so it is adequate. Typical optimum injection voltages will run from 1 to 10 volts for control grid injection, and 45 volts or so for screen or suppressor grid injection. There always are two signal frequencies which will combine with a given frequency to produce the same difference frequency. For example: assume a superheterodyne with its oscillator operating on a higher frequency than the signal, which is common practice in present superheterodynes, tuned to receive a signal at 14,100 kc. Assuming an i -f amplifier frequency of 450 kc., the mixer input circuit will be tuned to 14,100 kc., and the oscillator to 14,100 plus 450, or 14,550 kc. Now, a strong signal at the oscillator frequency plus the intermediate frequency (14,550 plus 450, or 15,000 kc.) will also give a difference frequency of 450 kc. in the mixer out put and will be heard also. Note that the image is always twice the intermediate frequency away from the desired signal. Images cause repeat points on the tuning dial. The only way that the image could be eliminated in this particular case would be to make the selectivity of the mixer input circuit, and any circuits preceding it, great enough so that the 15,000-kc. signal never reaches the mixer grid in sufficient amplitude to produce interference. For any particular intermediate frequency, image interference troubles become increasingly greater as the frequency to which the signal- frequency portion of the receiver is tuned is increased. This is due to the fact that the percentage difference between the desired frequency and the image frequency decreases as the receiver is tuned to a higher frequency. The ratio of strength between a signal at the image frequency and a signal at the frequency to which the receiver is tuned producing equal output is known as the image ratio. The higher this ratio, the better the receiver in regard to image- interference troubles. kith but a single tuned circuit between the mixer grid and the antenna, and with 400 -500 kc. i -f amplifiers, image ratios of 60 db and over are easily obtainable up to frequencies around 2000 kc. Above this frequency, greater selectivity in the mixer grid circuit through the use of additional tuned circuits between the mixer and the antenna is necessary if a good image ratio is to be maintained. Images 12 -5 Z -F Stages Since the necessLry tuned circuits between the mixer and the antenna can be combined with tubes to form r -f amplifier stages, the www.americanradiohistory.com Radio 212 Receiver Fundamentals PENTODE INPUT 6AB4, 6J6, 6J4, 12AT7 O GROUNDED-GRID C RADIO THE -i 70 Figure TYPICAL PENTODE R -F +120v. 8 AMPLIFIER STAGE CATHODE- COUPLED reduction of the effects of mixer noise and the increasing of the image ratio can be accomplished in a single section of the receiver. When incorporated in the receiver, this section is known simply as an r -/ amplifier; when it is a separate unit with a separate tuning control it is often known as a preselector. Either one or two stages are commonly used in the preselector or r -f amplifier. Some pre selectors use regeneration to obtain still greater amplification and selectivity. An r -f amplifier or preselector embodying more than two stages rarely ever is employed since two stages will ordinarily give adequate gain to override mixer noise. Generally speaking, atmospheric noise in the frequency range above 30 Mc. is quite low -so low, in fact, that the noise generated within the receiver itself is greater than the noise received on the antenna. Hence it is of the greatest importance that internally generated noise be held to a minimum in a receiver. At frequencies much above 300 Mc. there is not too much that can be done at the present state of the art in the direction of reducing receiver noise below that generated in the converter stage. But in the v -h -f range, between 30 and 300 Mc., the receiver noise factor in a well designed unit is determined by the characteristics of the first r -f stage. The usual v -h -f receiver, whether for communications or for FM or TV reception, uses a miniature pentode for the first r -f amplifier stage. The 6AK5 is the best of presently available types, with the 6CB6 and the 6DC6 closely approaching the 6AK5 in performance. But when gain in the first r -f stage is not so important, and the best noise factor must be obtained, the first r -f stage usually uses a triode. Shown in figure 9 are four commonly used types of triode r -f stages for use in the v -h -f range. The circuit at (A) uses few components and gives a moderate amount of gain with very low noise. It is most satisfactory when the first r -f stage is to be fed directly from a low- 6J6 +120 V LOW NOISE CASCODE Lo R -F Stages in the V -H -F Range +120 V. 6BK7,6B07AOR6BZ7 200V. Figure 9 TYPICAL TRIODE V -H -F R -F AMPLIFIER STAGES Triode r -f stages contribute the least amount of noise output for a given signal level, hence their frequent use in the v-h -f range. impedance coaxial transmission line. Figure 9 (B) gives somewhat more gain than (A), but requires an input matching circuit. The effective gain of this circuit is somewhat reduced when it is being used to amplify a broad band of frequencies since the effective Gm of the cathode -coupled dual tube is somewhat less www.americanradiohistory.com HANDBOOK The IA MC. Cascode R F. AMPLIFIER 10 MC. TUNABLE MIXER 213 455 KC. MC. TUNABLE Amplifier DULATOR FII% MIXER I.F. AMPLIFIER AMPLIFIER CRYSTAL OSCILLATOR VARIABLE OSCILLATOR AND AUDIO 3545 KC. 1 455 KC. MIXER 50 KC. I I nx I I MIXER I.I. AMPLIFIER l I L FIXED DEMODULATOR AMPLIFIER AND AUDIO II I I 1A4S5KC VARIABLE FIXED 11 OSCILLATOR OSCILLATOR I CONVENTIONAL COMMUNICATIONS RECEIVER 505 BC. I II HIGHLY SELECTIVE ACCESSORY II AMPLIFIER AND DEMODULATOR I F. (Q5'ER)I _JL Figure 10 TYPICAL DOUBLE -CONVERSION SUPERHETERODYNE RECEIVERS Illustrated at (A) is the basic circuit of a commercial double- conversion superheterodyne receiver. At (B) is illustrated the application of on accessory sharp i -f channel for obtaining improved selectivity from a conventional communications receiver through the use of the double-conversion principle. than half the taken alone. Gm of either of the two tubes The Cascode r -f amplifier, developed at the MIT Radiation Laboratory during World War II, is a low noise circuit employing a grounded cathode triode driving a grounded grid triode, as shown in figure 9C. The stage gain of such a circuit is about equal to that of a pentode tube, while the noise figure remains at the low level of a triode tube. Neutralization of the first triode tube is usually unnecessary below 50 Mc. Above this frequency, a definite improvement in the noise figure may be obtained Through the use of neutralization. The neutralizing coil, LN, should resonate at the operating frequency with the grid -plate capacity of the first triode tube. The 6B(27A and 6BZ7 tubes are designed for use in cascode circuits, and may be used to good advantage in the 144 Mc. and 220 Mc. amateur bands (figure 9D). For operation at higher frequencies, the 6A)4 tube is recommended. The Cascode Amplifier As previously mentioned, the use of a higher intermediate frequency will also improve the image Double Conversion ratio, at the expense of i -f selectivity, by placing the desired signal and the image farther apart. To give both good image ratio at the higher frequencies and good selectivity in the i -f amplifier, a system known as double conversion is sometimes employed. In this system, the incoming signal is first converted to a rather high intermediate frequency, and then amplified and again converted, this time to a much lower frequency. The first intermediate frequency supplies the necessary wide separation between the image and the desired signal, while the second one supplies the bulk of the i -f selectivity. The double -conversion system, as illustrated in figure 10, is receiving two general types of application at the present time. The first application is for the purpose of attaining extremely good stability in a communications receiver through the use of crystal control of the first oscillator. In such an arrangement, as used in several types of Collins receivers, the first oscillator is crystal controlled and is followed by a tunable i -f amplifier which then is followed by a mixer stage and a fixed-tuned i -f a m p l i f i e r on a touch lower frequency. Through such a circuit arrangement the sta- bility of the complete receiver is equal to the www.americanradiohistory.com 214 Radio Receiver THE Fundamentals RADIO stability of the oscillator which feeds the second mixer, while the selectivity is determined by the bandwidth of the second, fixed i -f am- plifier. The second common application of the double- conversion principle is for the purpose of obtaining a very high degree of selectivity in the complete communications receiver. In this type of application, as illustrated in figure 10 (B), a conventional communications receiver is modified in such a manner that its normal i -f amplifier (which usually is in the 450 to 915 kc. range) instead of being fed to a demodulator and then to the audio system, is alternatively fed to a fixed -tune mixer stage and then into a much lower intermediate frequency amplifier before the signal is demodulated and fed to the audio system. The accessory i -f amplifier system (sometimes called a Q5'er) normally is operated on a frequency of 175 kc., 85 kc., or 50 kc. 12 -6 Signal- Frequency Tuned Circuits The signal- frequency tuned circuits in high frequency superheterodynes and tuned radio frequency types of receivers consist of coils of either the solenoid or universal -wound types shunted by variable capacitors. It is in these tuned circuits that the causes of success or failure of a receiver often lie. The universal wound type coils usually are used at frequencies below 2000 kc.; above this frequency the single-layer solenoid type of coil is more satisfactory. The two factors of greatest significance in determining the gain per -stage and selectivity, respectively, of a tuned amplifier are tuned- circuit impedance and tuned -circuit Q. Since the resistance of modern capacitors is low at ordinary frequencies, the resistance usually can be considered to be concentrated in the coil. The resistance to be considered in making Q determinations is the r -f resistance, not the d -c resistance of the wire in the coil. The latter ordinarily is low enough that it may be neglected. The increase in r -f resistance over d -c resistance primarily is due to skin effect and is influenced by such factors as wire size and type, and the proximity of metallic objects or poor insulators, such as coil forms with high losses. Higher values of Q lead to better selectivity and increased r -f voltage across the tuned circuit. The increase in voltage is due to an increase in the circuit impedance with the higher values of Q. Impedance and Q R F C INPUT Figure 11 ILLUSTRATING "COMMON POINT" BY- PASSING To reduce the detrimental effects of cathode circuit inductance in v -h -f stages, all by -pass capacitors should be returned to the cathode terminal at the socket. Tubes with two cathode leads can give improved performance if the grid return is made to one cathode terminal while the plate and screen by -pass returns are made to the cathode terminal which is connected to the suppressor within the tube. Frequently it is possible to secure an increase in impedance in a resonant circuit, and consequently an increase in gain from an amplifier stage, by increasing the reactance through the use of larger coils and smaller tuning capacitors (higher L/C ratio). Another factor which influences the operation of tuned circuits is the input resistance of the tubes placed across these circuits. At broadcast frequencies, the input resistance of most conventional r -f amplifier tubes is high enough so that it is not bothersome. But as the frequency is increased, the input resistance becomes lower and lower, until it ultimately reaches a value so low that no amplification can be obtained from the r -f stage. The two contributing factors to the decrease in input resistance with increasing frequency are the transit time required by an electron traveling between the cathode and grid, and the inductance of the cathode lead common to both the plate and grid circuits. As the frequency becomes higher, the transit time can become an appreciable portion of the time required by an r -f cycle of the signal voltage, and current will actually flow into the grid. The result of this effect is similar to that which would be obtained by placing a resistance between the tube's grid and cathode. Input Resistance Because the oscillator in a superheterodyne operate s "offset" from the other front end circuits, it is necessary to make special provisions to allow the oscillator to track Superheterodyne Tracking when similar tuning capacitor sections are www.americanradiohistory.com Tuning Circuits HANDBOOK 215 MIXER PADDING CAPACITOR TUNING CAPACITOR OSCILLATOR SERIES TRACKING CAPACITOR Figure 13 BANDSPREAD CIRCUITS Parallel bandspread is illustrated at (A) and (B), series bandspread at (C), and tq,ped.coil band- Figure 12 SERIES TRACKING EMPLOYED IN THE H -F OSCILLATOR OF A SUPERHETERODYNE The series tracking capacitor permits the use of identical gangs in a ganged capacitor, since the tracking capacitor slows down the rate of frequency change in the oscillator so that a constant difference in frequency between the oscillator and the r -f stage (equal to the i -f amplifier frequency) may be maintained. ganged. The usual method of obtaining good tracking is to operate the oscillator on the high- frequency side of the mixer and use a series tracking capacitor to slow down the tuning rate of the oscillator. The oscillator tuning rate must be slower because it covers a smaller range than does the mixer when both are expressed as a percentage of frequency. At frequencies above 7000 kc. and with ordinary intermediate frequencies, the difference in percentage between the two tuning ranges is so small that it may be disregarded in receivers designed to cover only a small range, such as an amateur band. A mixer and oscillator tuning arrangement in which a series tracking capacitor is provided is shown in figure 12. The value of the tracking capacitor varies considerably with different intermediate frequencies and tuning ranges, capacitances as low as .0001 pfd. being used at the lower tuning -range frequencies, and values up to .01 µfd. being used at the higher frequencies. Superheterodyne receivers designed to cover only a single frequency range, such as the standard broadcast band, sometimes obtain tracking between the oscillator and the r -f circuits by cutting the variable plates of the oscillator tuning section to a different shape from those used to tune the r -f stages. frequency to which a receiver responds may be varied by changing the size of either the coils or the capacitors in the tuning circuits, or both. In short -wave receivers Frequency Range Selection The spread at (D), combination of both methods is usually employed, the coils being changed from one band to another, and variable capacitors being used to tune the receiver across each band. In practical receivers, coils may be changed by one of two methods: a switch, controllable from the panel, may be used to switch coils of different sizes into the tuning circuits or, alternatively, coils of different sizes may be plugged manually into the receiver, the connection into the tuning circuits being made by suitable plugs on the coils. Where there are several plug -in cods for each band, they are sometimes arranged to a single mounting strip, allowing them all to be plugged in simultaneously. a In receivers using large tuning capacitors to cover the shortwave spectrum with a minimum of coils, tuning is likely to be quite difficult, owing to the large frequency range covered by a small rotation of the variable capacitors. To alleviate this condition, some method of slowing down the tuning rate, or bandspread ing, must be used. Bandspread Tuning Quantitatively, bandspread is usually designated as being inversely proportional to the range covered. Thus, a large amount of bandspread indicates that a small frequency range is covered by the bandspread control. Conversely, a small amount of bandspread is taken to mean that a large frequency range is covered by the bandspread dial. Types of Bandspread Bandspreading systems are of two general types: electrical and mechanical. Mechanical systems are exemplified by high -ratio dials in which the tuning capacitors rotate much more slowly www.americanradiohistory.com 216 Radio Receiver THE Fundamentals than the dial knob. In this system, there is often a separate scale or pointer either connected or geared to the dial knob to facilitate accurate dial readings. However, there is a practical limit to the amount of mechanical bandspread which can be obtained in a dial and capacitor before the speed- reduction unit and capacitor bearings become prohibitively expensive. Hence, most receivers employ a combination of electrical and mechanical band spread. In such a system, a moderate reduction in the tuning rate is obtained in the dial, and the rest of the reduction obtained by elec- trical bandspreading. In this book and in other radio literature, mention is sometimes made of stray or circuit capacitance. This capacitance is in the usual sense defined as the capacitance remaining across a coil when all the tuning, bandspread, and padding capacitors across the circuit are at their minimum capacitance setting. Circuit capacitance can be attributed to two general sources. One source is that due to the input and output capacitance of the tube when its cathode is heated. The input capacitance varies somewhat from the static value when the tube is in actual operation. Such factors as plate load impedance, grid bias, and frequency will cause a change in input capacitance. However, in all except the extremely high -transconductance tubes, the published measured input capacitance is reasonably close to the effective value when the tube is used within its recommended frequency range. But in the high -transconductance types the effective capacitance will vary considerably from the published figures as operating conditions are changed. The second source of circuit capacitance, and that which is more easily controllable, is that contributed by the minimum capacitance of the variable capacitors across the circuit and that due to capacitance between the wiring and ground. In well -designed high -frequency receivers, every effort is made to keep this portion of the circuit capacitance at a minimum since a large capacitance reduces the tuning range available with a given coil and prevents a good L/C ratio, and consequently a high- impedance tuned circuit, from being obtained. A good percentage of stray circuit capacitance is due also to distributed capacitance of the coil and capacitance between wiring Stray Circuit Capacitance points and chassis. Typical values of circuit capacitance may run from 10 to 75 µpfd. in high- frequency re- ceivers, the first figure representing concentric-line receivers with acorn or miniature tubes and extremely small tuning capacitors, RADIO latter representing all -wave sets with bandswitching, large tuning capacitors, and conventional tubes. and the 12 -7 I -F Tuned Circuits I -f amplifiers usually employ bandpass circuits of some sort. A bandpass circuit is exactly what the name implies -a circuit for passing a band of frequencies. Bandpass arrange- ments can be designed for almost any degree of selectivity, the type used in any particular case depending upon the ultimate application of the amplifier. I.F Transformers Intermediate frequency trans formers ordinarily consist of two or more tuned circuits and some method of coupling the tuned circuits together. Some representative arrangements are shown in figure 14. The circuit shown at A is the conventional i -f transformer, with the coupling, M, between the tuned circuits being provided by inductive coupling from one coil to the other. As the coupling is increased, the selectivity curve becomes less peaked, and when a condition known as critical coupling is reached, the top of the curve begins to flatten out. When the coupling is increased still more, a dip occurs in the top of the curve. The windings for this type of i -f transformer, as well as most others, nearly always consist of small, flat universal -wound pies mounted either on a piece of dowel to provide an air core or on powdered -iron for iron core i-f transformers. The iron -core transformers generally have somewhat more gain and better selectivity than equivalent air-core units. The circuits shown at figure 14 -B and C are quite similar. Their only difference is the type of mutual coupling used, an inductance being used at B and a capacitance at C. The operation of both circuits is similar. Three resonant circuits are formed by the components. In B, for example, one resonant circuit is formed by C1, C, and L2 all in series. The frequency of this resonant circuit is just the same as that of a single one of the coils and capacitors, since the coils and capacitors are similar in both sides of the circuit, and the resonant frequency of the two capacitors and the two coils all in series is the same as that of a single coil and capacitor. The second resonant frequency of the complete circuit is determined by the characteristics of each half of the circuit containing the mutual coupling device. In B, this second frequency will be lower than the first, since the resonant frequency of C, and the inductance, M, or L2, C, and M is lower than that of a single coil and capaci- L L www.americanradiohistory.com HANDBOOK I tor, due to the inductance of M being added to the circuit. The opposite effect takes place at figure 14 -C, where the common coupling impedance is a capacitor. Thus, at C the second resonant frequency is higher than the first. In either case, however, the circuit has two resonant frequencies, resulting in a flat- topped selectivity curve. The width of the top of the curve is controlled by the reactance of the mutual coupling component. As this reactance is increased (inductance made greater, capacitance made smaller), the two resonant frequencies become further apart and the curve is broadened. In the circuit of figure 14 -D, there is inductive coupling between the center coil and each of the outer coils. The result of this arrangement is that the center coil acts as a sharply tuned coupler between the other two. A signal somewhat off the resonant frequency of the transformer will not induce as much current in the center coil as will a signal of the correct frequency. When a smaller current is induced in the center coil, it in turn transfers a still smaller current to the output coil. The effective coupling between the outer coils increases as the resonant frequency is approached, and remains nearly constant over a small range and then decreases again as the resonant band is passed. Another very satisfactory bandpass arrangement, which gives a very straight- sided, flat topped curve, is the negative- mutual arrangement shown at figure 14 -E. Energy is transferred between the input and output circuits in this arrangement by both the negative- mutual coils, M, and the common capacitive reactance, C. The negative- mutual coils are interwound on the same form, and connected backward. Transformers usually are made tunable over a small range to permit accurate alignment in the circuit in which they are employed. This is accomplished either by means of a variable capacitor across a fixed inductance, or by means of a fixed capacitor across a variable inductance. The former usually employ either mica -compression capacitor (designated a "mica tuned "), or a small air dielectric variable capacitor (designated "air tuned"). Those which use a fixed capacitor usually employ a powdered iron core on a threaded rod to vary the inductance, and are known as "permea- bility tuned." It is obvious that to pass modulation sidebands and to allow for slight drifting of the transmitter carrier frequency and the receiver local oscillator, the i -f amplifier must pass not a single frequency but a band of frequencies. The width of this pass band, usually 5 to 8 kc. at maximum Shape Factor -F M Amplifiers 217 M E Figure 14 -F AMPLIFIER COUPLING ARRANGEMENTS The interstoge coupling arrangements illustrated above give a better shape factor (more straight sided selectivity curve) than would the some number of tuned circuits coupled by means of tubes. I width in a good communications receiver, is known as the pass band, and is arbitrarily taken as the width between the two frequencies at which the response is attenuated 6 db, or is "6 db down." However, it is apparent that to discriminate against an interfering signal which is stronger than the desired signal, much more than 6 db attenuation is required. The attenuation arbitrarily taken to indicate adequate discrimination against an interfering signal is 60 db. www.americanradiohistory.com 218 Radio Receiver THE Fundamentals L R C Figure RADIO 16 ELECTRICAL EQUIVALENT OF QUARTZ FILTER CRYSTAL The crystal is equivalent to o very large value of inductance in series with small values of capacitance and resistance, with a larger though still small value of capacitance across the whole circuit Yrepresenting holder capacitance plus stray capacitances). Figure 15 -F PASS BAND OF TYPICAL COMMUNICATIONS RECEIVER I It is apparent that it is desirable to have bandwidth at 60 db down as narrow as possible, but it must be done without making the pass band (6 db points) too narrow for satisfactory reception of the desired signal. The figure of merit used to show the ratio of bandwidth at 6 db down to that at 60 db down is designated shape factor. The ideal i -f curve, a rectangle, would have a shape factor of 1.0. The i -f shape factor in typical communications receivers runs from 3.0 to 5.5. The most practicable method of obtaining a low shape factor for a given number of tuned circuits is to employ them in pairs, as in figure 14 -A, adjusted to critical coupling (the value at which two resonance points just begin to become apparent). If this gives too sharp a "nose" or pass band, then coils of lower Q should be employed, with the coupling maintained at the critical value. As the Q is lowered, closer coupling will be required for critical coupling. Conversely if the pass band is too broad, coils of higher Q should be employed, the coupling being maintained at critical. If the pass band is made more narrow by using looser coupling instead of raising the Q and main taninig critical coupling, the shape factor will not be as good. The pass band will not be much narrower for several pairs of identical, critically coupled tuned circuits than for a single pair. However, the shape factor will be greatly improved as each additional pair is added, up to about 5 pairs, beyond which the improvement for each additional pair is not significant. Commercially available communications receivers of the good quality normally employ 3 or 4 double tuned transformers with coupling adjusted to critical or slightly less. The pass band of a typical communication receiver having a 455 kc. i -f amplifier is shown in figure 15. "Miller Effect" As mentioned previously, the dynamic input capacitance of a tube varies slightly with bias. As a -v-c voltage normally is applied to i -f tubes for radiotelephony reception, the effective grid-cathode capacitance varies as the signal strength varies, which produces the same effect as slight detuning of the i -f transformer. This effect is known as "Miller effect," and can be minimized to the extent that it is not troublesome either by using a fairly low L/C ratio in the transformers or by incorporating a small amount of degenerative feedback, the latter being most easily accomplished by leaving part of the cathode resistor unbypassed for r.f. Crystal Filters The pass band of an intermediate frequency amplifier may be made very narrow through the use of a piezoelectric filter crystal employed as a series resonant circuit in a bridge arrangement known as a crystal filter. The shape factor is quite poor, as would be expected when the selectivity is obtained from the equivalent of a single tuned circuit, but the very narrow pass band obtainable as a result of the extremely high Q of the crystal makes the crystal filter useful for c -w telegraphy reception. The pass band of a 455 kc. crystal filter may be made as narrow as 50 cycles, while the narrowest pass band that can be obtained with a 455 kc. tuned circuit of practicable dimensions is about 5 kc. The electrical equivalent of a filter crystal is shown in figure 16. For a given frequency, L is very high, C very low, and R (assuming www.americanradiohistory.com Filters Crystal HANDBOOK 219 CRYSTAL E SELECTIVITY CONTROL PHASING CONTROL Figure For a Figure 17 EQUIVALENT OF CRYSTAL FILTER CIRCUIT given voltage out of the generator, the volt- age developed across Z1 depends upon the ratio of the impedance of X to the sum of the impedances of Z and Z1. Because of the high of the crystal, its impedance changes rapidly with changes in Q frequency. a good crystal of high Q) is very low. Capacitance C, represents the shunt capacitance of the electrodes, plus the crystal holder and wiring, and is many times the capacitance of C. This makes the crystal act as a parallel resonant circuit with a frequency only slightly higher than that of its frequency of series resonance. For crystal filter use it is the series resonant characteristic that we are primarily interested in. The electrical equivalent of the basic crystal filter circuit is shown in figure 17. If the impedance of Z plus Z, is low compared to the impedance of the crystal X at resonance, then the current flowing through and the voltage developed across it, will be almost in inverse proportion to the impedance of X, which has a very sharp resonance curve. If the impedance of Z plus Z, is made high compared to the resonant impedance of X, then there will be no appreciable drop in voltage across Z, as the frequency departs from the resonant frequency of X until the point is reached where the impedance of X approaches that of Z plus Z,. This has the effect of broadening out the curve of frequency versus voltage developed across which is another way of saying that the selectivity of the crystal filter (but not the crystal proper) has been reduced. In practicable filter circuits the impedances Z and Z, usually are represented by some form of tuned circuit, but the basic principle of operation is the same. Z Z Fractical Filters It is necessary to balance out the capacitance across the crystal holder (C in figure 16) to prevent bypassing around the crystal undesired signals off the crystal resonant frequency. The balancing is done by a phasing circuit which takes out -of-phase voltage from a balanced in- 18 TYPICAL CRYSTAL FILTER CIRCUIT put circuit and passes it to the output side of the crystal in proper phase to neutralize that passed through the holder capacitance. A rep- resentative practical filter arrangement is shown in figure 18. The balanced input circuit may be obtained either through the use of a split- stator capacitor as shown, or by the use of a center -tapped input coil. Variable- Selec- circuit of figure 18, the selectivity is minimum with the crystal input circuit tuned to resonance, since at resonance the impedance of the tuned circuit is maximum. As the input circuit is detuned from resonance, however, the impedance decreases, and the selectivity becomes greater. In this circuit, the output from the crystal filter is tapped down on the i -f stage grid winding to provide a low value of series impedance in the output circuit. It will be recalled that for maximum selectivity, the total impedance in series with the crystal (both input and output circuits) must be low. If one is made low and the other is made variable, then the selectivity may be tivity Filters In the varied at will from sharp to broad. The circuit shown in figure 19 also achieves variable selectivity by adding a variable impedance in series with the crystal circuit. In this case, the variable impedance is in series with the crystal output circuit. The impedance of the output circuit is varied by varying the Q. As the Q is reduced (by adding resistance in series with the coil), the impedance decreases and the selectivity becomes greater. The input circuit impedance is made low by using a non -resonant secondary on the input transformer. A variation of the circuit shown at figure 19 consists of placing the variable resistance across the coil and capacitor, rather than in series with them. The result of adding the resistor is a reduction of the output impedance, and an increase in selectivity. The circuit behaves oppositely to that of figure 19, however; as the resistance is lowered the selectivity becomes greater. Still another variation of figure 19 is to use the tuning capacitor across the output coil to vary the output impedance. www.americanradiohistory.com 220 Radio Receiver Fundamentals RADIO THE CRYSTwL SELECTIVITY CONTROL Figure r CRYSTAL NOTCH +I +2 +3 z 3 3S 19 VARIABLE SELECTIVITY CRYSTAL FILTER O This circuit permits of a greater control of selectivity than does the circuit of figure 16, and does not require a split- stator variable capacitor. m 0 40 J V CI -3 -I -2 so 455 + KC Figure 20 -F PASS BAND OF TYPICAL CRYSTAL FILTER COMMUNICATIONS RECEIVER As the output circuit is detuned from resonance, its impedance is lowered, and the selectivity increases. Sometimes a set of fixed capacitors and a multipoint switch are used to give step -by -step variation of the output circuit tuning, and thus of the crystal filter selectivity. As previously discussed, a filter crystal has both a resonant(series resonant) and an anti -resonant (parallel resonant) frequency, the impedance of the crystal being quite low at the former frequency, and quite high at the latter frequency. The anti- resonant frequency is just slightly higher than the resonant frequency, the difference depending upon the effective shunt capacitance of the filter crystal and holder. As adjustment of the phasing capacitor controls the effective shunt capacitance of the crystal, it is possible to vary the anti -resonant frequency of the crystal slightly without unbalancing the circuit sufficiently to let undesired signals leak through the shunt capacitance in appreciable amplitude. At the exact anti -resonant frequency of the crystal the attenuation is exceedingly high, because of the high impedance of the crystal at this frequency. This is called the rejection notch, and can be utilized virtually to eliminate the heterodyne image or repeat tuning of c -w signals. The beat frequency oscillator can be so adjusted and the phasing capacitor so adjusted that the desired beat note is of such a pitch that the image (the same audio note on the other side of zero beat) falls in the rejection notch and is inaudible. The receiver then is said to be adjusted for single -signal Rejection Notch operation. The rejection notch sometimes can be employed to reduce interference from an undesired phone signal which is very close in frequency to a desired phone signal. The filter is adjusted to "broad" so as to permit tele- I phony reception, and the receiver tuned so that the carrier frequency of the undesired signal falls in the rejection notch. The modulation sidebands of the undesired signal still will come through, but the carrier heterodyne will be effectively eliminated and interference greatly reduced. A typical crystal selectivity curve for a communications receiver is shown in figure 20. Crystal Filter Considerations A crystal filter, especially when adjusted for single sig- nal reception, greatly reduces interference and background noise, the latter feature permitting signals to be copied that would ordinarily be too weak to be heard above the background hiss. However, when the filter is adjusted for maximum selectivity, the pass band is so narrow that the received signal must have a high order of stability in order to stay within the pass band. Likewise, the local oscillator in the receiver must be highly stable, or constant retuning will be required. Another effect that will be noticed with the filter adjusted too "sharp" is a tendency for code characters to produce a ringing sound, and have a hangover or "tails." This effect limits the code speed that can be copied satisfactorily when the filter is adjusted for extreme selectivity. The Collins Mechanical Fil ter (figure 21) is a new concept in the field of selectivity. It is an electro- mechanical bandpass filter about half the size of a cigarette package. As shown in figure 22, it consists of an input transducer, a resonant mechanical secThe Mechanical Filter www.americanradiohistory.com Collins Mechanical Filter HANDBOOK tion comprised of a number of metal discs, and an output transducer. The frequency characteristics of the resonant mechanical section provide the almost rectangular selectivity curves shown in figure 23. The input and output transducers serve only as electrical to mechanical coupling devices and do not affect the selectivity characteristics which are determined by the metal discs. An electrical signal applied to the input terminals is converted into a mechanical vibration at the input transducer by means of magnetostriction. This mechanical vibration travels through the resonant mechanical section to the output transducer, where it is converted by magnetostriction to an electrical signal which appears at the output terminals. In order to provide the most efficient electromechanical coupling, a small magnet in the mounting above each transducer applies a magnetic bias to the nickel transducer core. The electrical impulses then add to or subtract from this magnetic bias, causing vibration of the filter elements that corresponds to the exciting signal. There is no mechanical motion except for the imperceptible vibration of the metal discs. Magnetostrictively -driven mechanical filters have several advantages over electrical equivalents. In the region from 100 kc. to 500 kc., the mechanical elements are extremely small, and a mechanical filter having better selectivity than the best of conventional i -f systems may be enclosed in a package smaller than one i -f transformer. Since mechanical elements with Q's of 5000 or more are readily obtainable, mechanical filters may be designed in accordance with the theory for lossless elements. This permits filter characteristics that are unobtainable with electrical circuits because of the relatively high losses in electrical elements as compared with the mechanical elements used in the 221 c. Figure 21 COLLINS MECHANICAL FILTERS The Collins Mechanical Filter is an electro- mechanical bandpass filter which surpasses, in one small unit, the selectivity of conventional, space-consuming filters. At the left is the miniaturized filter, less than 2!4' long. Type H is next, and two horizontal mounting types are at right. For exploded view of Collins Mechanical Filter, see figure 46. The frequency characteristics of the mechanical filter are permanent, and no adjustment is required or is possible. The filter is enclosed in a hermetically sealed case. In order to realize full benefit from the mechanical filter's selectivity characteristics, it is necessary to provide shielding between the external input and output circuits, capable of reducing transfer of energy external to the o filters. ONE SUPPORTING DISC AT EACH END RESONANT MECHANICAL SECTION (0 RESONANT DISCS) ill COUPLING RODS DIAS MAGNET iì%U MAGNETOSTRICTIVE DRIVING ROD RANSDUCER COIL ELECTRICAL SIGNAL (INPUT OR OUTPUT) ELECTRICAL SIGNAL (INPUT OR OUTPUT) Figure 22 MECHANICAL FILTER FUNCTIONAL DIAGRAM Figure 23 Selectivity curves of 4554c. mechanical filters with nominal 0.8 -%c. (dotted line) and 3.1 -kc. (solid line) bandwidth at -6 db. www.americanradiohistory.com 222 Radio Receiver THE Fundamentals I VERY SMALL 65J7 I.F. STAGE RADIO DET. AUDIO //1 Por I-rlp 2Np I DETECTOR BY J pA I.F STAGE Figure 24 A GRID LEAK DETECTOR DET. VARIABLE -OUTPUT B -F -O CIRCUIT beat- frequency oscillator whose output is con- trollable is of considerable assistance in copying c-w signals over a wide ronge of levels, and such a control is almost a necessity for satisfactory copying of single -sideband radiophone signals. AUDIO filter by a minimum value of 100 db. If the input circuit is allowed to couple energy into the output circuit external to the filter, the excellent skirt selectivity will deteriorate and the passband characteristics will be distorted. As with almost any mechanically resonant circuit, elements of the mechanical filter have multiple resonances. These result in spurious modes of transmission through the filter and produce minor passbands at frequencies on other sides of the primary passband. Design of the filter reduces these sub -bands to a low level and removes them from the immediate area of the major passband. Two conventional i -f transformers supply increased attenuation to these spurious responses, and are sufficient to reduce them to an insignificant level. The beat -frequency oscillator, usually called the b.J.o., is a necessary adjunct for reception of c -w telegraph signals on superheterodynes which have no other provision for obtaining modulation of an incoming c -w telegraphy signal. The oscillator is coupled into or just ahead of second detector circuit and supplies a signal of nearly the same frequency as that of the desired signal from the i -f amplifier. If the i -f amplifier is tuned to 455 kc., for example, the b.f.o. is tuned to approximately 454 or 456 kc. to produce an audible (1000 cycle) beat note in the output of the second detector of the receiver. The carrier signal itself is, of course, inaudible. The b.f.o. is not used for voice reception, except as an aid in searching for weak stations. The b -f -o input to the second detector need only be sufficient to give a good beat note on an average signal. Too much coupling into the second detector will give an excessively high hiss level, masking weak signals by the high noise background. Figure 24 shows a method of manually ad- OB DIODE DETECTOR I.F. STAGE DET. AUDIO © PLATE DETECTOR I.F. STAGE DET. Beat- Frequency Oscillators OD INFINITE IMPEDANCE DETECTOR Figure 25 TYPICAL CIRCUITS FOR GRID -LEAK, DIODE, PLATE AND INFINITE IMPEDANCE DETECTOR STAGES justing the b -f -o output to correspond with the strength of received signals. This type of variable b -f -o output control is a useful adjunct to any superheterodyne, since it allows sufficient b-f-o output to be obtained to beat with strong signals or to allow single - sideband reception and at the same time permits the b -f -o output, and consequently the hiss, to be reduced when attempting to receive weak signals. The circuit shown is somewhat better than those in which one of the electrode volt- www.americanradiohistory.com HANDBOOK Detector Circuits TYPICAL A -V -C 223 Figure 26 CIRCUIT USING A DOUBLE DIODE Any of the small dual-diode tubes may be used in this circuit. Or, if desired, a duo- diodetriode may be used, with the triode acting as the first audio stage. The left-hand diode serves as the detector, while the right -hand side acts as the a-v -c rectifier. The use of separate diodes for detector and o-v -c reduces distortion when receiving an AM signal with a high modulation percentage. b -f-o tube is changed, as the latter circuits usually change the frequency of the ages on the b.f.o. at the same time they change the strength, making it necessary to reset the trimmer each time the output is adjusted. The b.f.o. usually is provided with a small trimmer which is adjustable from the front panel to permit adjustment over a range of 5 or 10 kc. For single -signal reception the b.f.o. always is adjusted to the high- frequency side, in order to permit placing the heterodyne image in the rejection notch. In order to reduce the b -f -o signal output voltage to a reasonable level which will prevent blocking the second detector, the signal voltage is delivered through a low- capacitance (high -reactance) capacitor having a value of 1 to 2 fgifd. Care must be taken with the b.f.o. to prevent harmonics of the oscillator from being picked up at multiples of the b -f-o frequency. The complete b.f.o. together with the coupling circuits to the second detector, should be thoroughly shielded to prevent pickup of the harmonics by the input end of the receiver. If b-f-o harmonics still have a tendency to give trouble after complete shielding and isolation of the b -f-o circuit has been accomplished, the passage of these harmonics from the b-f-o circuit to the rest of the receiver can be stopped through the use of a low -pass filter in the lead between the output of the b -f -o circuit and the point on the receiver where the b-f-o signal is to be injected. 12 -8 Detector, Audio, and Control Circuits Detectors Second detectors for use in superheterodynes are usually of the diode, plate, or infinite-impedance types. Occasionally, grid-leak detectors are used in receivers using one i-f stage or none at all, in which case the second detector usually is made regenerative. Diodes are the most popular second detectors because they allow a simple method of obtaining automatic volume control to be used. Diodes load the tuned circuit to which they are connected, however, and thus reduce the selectivity slightly. Special i -f transformers are used for the purpose of providing a low impedance input circuit to the diode detector. Typical circuits for grid -leak, diode, plate and infinite -impedance detectors are shown in figure 25. The elements of an automatic volume control (a.v.c.) system are shown in figure 26. A dual -diode tube is used as a combination diode detector and a -v -c rectifier. The left hand diode operates as a simple rectifier in the manner described earlier in this chapter. Audio voltage, superimposed on a d -c voltage, appears across the 500,000 -ohm potentiometer (the volume control) and the .0001 -µfd. capacitor, and is passed on to the audio amplifier. The right -hand diode receives signal voltage directly from the primary of the last i -f amplifier, and acts as the a -v -c rectifier. The pulsating d-c voltage across the 1- megohm a.v.c.diode load resistor is filtered by a 500,000 -ohm resistor and a .05-pfd. capacitor, and applied as bias to the grids of the r -f and i -f amplifier tubes; an increase or decrease in signal strength will cause a corresponding increase or decrease in a-v -c bias voltage, and thus the gain of the receiver is automatically adjusted to compensate for changes in signal strength. Automatic Volurne Control www.americanradiohistory.com 224 Radio A -C Loading of Second Detector Receiver THE Fundamentals By disassociating the a.v.c. and detecting functions through using separate diodes, as shown, most of the ill effects of a-c shunt loading on the detector diode are avoided. This type of loading causes serious distortion, and the additional components required to eliminate it are well worth their cost. Even with the circuit shown, a -c loading can occur unless a very high (5 megohms, or more) value of grid resistor is used in the following audio amplifier stage. A.V.C. in IF IF RF RADIO RFoR IF In receivers having a beat frequency oscillator for the reception of radiotelegraph signals, the use of a.v.c. can result in a great loss in sensitivity when the b.f.o. is switched on. This is because the beat oscillator output acts exactly like a strong received signal, and causes the a -v -c circuit to put high bias on the r -f and i -f stages, thus greatly reducing the receiver's sensitivity. Due to the above effect, it is necessary to provide a method of making the a -v -c circuit inoperative when the b.f.o. is being used. The simplest method of eliminating the a -v -c action is to short the a -v-c line to ground when the b.f.o. is turned on. A two -circuit switch may be used for the dual purpose of turning on the beat oscillator and shorting out the a.v.c. if desired. B-F-O- Equipped Receivers Visual means for determining whether or not the receiver is properly tuned, as well as an indication of the relative signal strength, are both provided by means of tuning indicators (S meters) of the meter or vacuum -tube type. A d -c milliammeter can be connected in the plate supply circuit of one or more r -f or i -f amplifiers, as shown in figure 27A, so that the change in plate current, due to the action of the a -v -c voltage, will be indicated on the instrument. The d -c instrument MA should have a full -scale reading approximately equal to the total plate current taken by the stage or stages whose plate current passes through the instrument. The value of this current can be estimated by assuming a plate current on each stage (with no signal input to the receiver) of Signal Strength R Fon I F +70 6U5/6G5 OR 6E5 TO A V C Indicators about 6 ma. However, it will be found to be more satisfactory to measure the actual plate current on the stages with a milliammeter of perhaps 0 -100 ma. full scale before purchasing an instrument for use as an S meter. The 50 -ohm potentiometer shown in the drawing is used to adjust the meter reading to full scale with no signal input to the receiver. When an ordinary meter is used in the plate circuit of a stage, for the purpose of indicating signal strength, the meter reads backwards t250v Figure 27 SIGNAL -STRENGTH -METER CIRCUITS Shown above are four circuits for obtaining a signal- strength reading which is a function of incoming carrier amplitude. The circuits are discussed lin the accompanying text. with respect to strength. This is because increased a -v -c bias on stronger signals causes lower plate current through the meter. For this reason, special meters which indicate zero at the right -hand end of the scale are often used for signal strength indicators in commercial receivers using this type of circuit. Alternatively, the meter may be mounted upside down, so that the needle moves toward the right with increased strength. The circuit of figure 27B can frequently be used to advantage in a receiver where the cathode of one of the r-f or i -f amplifier stages runs directly to ground through the cathode bias resistor instead of running through a cath- www.americanradiohistory.com HANDBOOK ode -voltage gain control. In this case a 0 -1 d -c milliammeter in conjunction with a resistor from 1000 to 3000 ohms can be used as shown as a signal- strength meter. With this circuit the meter will read backwards with increasing signal strength as in the circuit previously discussed. Figure 27C is the circuit of a forward-readingS meter as is often used in communications receivers. The instrument is used in an unbalanced bridge circuit with the d -c plate resistance of one i -f tube as one leg of the bridge and with resistors for the other three legs. The value of the resistor R must be determined by trial and error and will be somewhere in the vicinity of 50,000 ohms. Sometimes the screen circuits of the r -f and i -f stages are taken from this point along with the screen-circuit voltage divider. Electron -ray tubes (sometimes called "magic eyes") can also be used as indicators of relative signal strength in a circuit similar to that shown in figure 27D. A 6U5/6G5 tube should be used where the a-v -c voltage will be from 5 to 20 volts and a type 6E5 tube should be used when the a-v-c voltage will run from 2 to 8 volts. amplifiers are employed in nearly all radio receivers. The audio amplifier stage or stages are usually of the Class A type, although Class AB push -pull stages are used in some receivers. The purpose of the audio amplifier is to bring the relatively weak signal from the detector up to a strength sufficient to operate a pair of headphones or a loud speaker. Either triodes, pentodes, or beam tetrodes may be used, the pentodes and beam tetrodes usually giving greater output. In some receivers, particularly those employing grid leak detection, it is possible to operate the headphones directly from the detector, without audio amplification. In such receivers, a single audio stage with a beam tetrode or pentode tube is ordinarily used to drive the loud speaker. Most communications receivers, either home constructed or factory -made, have a single ended beam tetrode (such as a 6L6 or 6V6) or pentode (6F6 or 6K6 -GT) in the audio output stage feeding the loudspeaker. If precautions are not taken such a stage will actually bring about a decrease in the effective signal -tonoise ratio of the receiver due to the rising high- frequency characteristic of such a stage when feeding a loud -speaker. One way of improving this condition is to place a mica or paper capacitor of approximately 0.003 pfd. capacitance across the primary of the output transformer. The use of a capacitor in this manner tends to make the load impedance seen by the plate of the output tube more constant Audio Ampl if iers Audio Noise Suppression 225 over the audio -frequency range. The speaker and transformer will tend to present a rising impedance to the tube as the frequency increases, and the parallel capacitor will tend to make the total impedance more constant since it will tend to present a decreasing impedance with increasing audio frequency. A still better way of improving the frequency characteristic of the output stage, and at the same time reducing the harmonic distortion, is to use shunt feedback from the plate of the output tube to the plate of a tube such as a 6SJ7 acting as an audio amplifier stage ahead of the output stage. Noise Suppression 12-9 The problem of noise suppression confronts the listener who is located in places where interference from power lines, electrical appliances, and automobile ignition systems is troublesome. This noise is often of such intensity as to swamp out signals from desired stations. There are two principal methods for reducing this noise: (1) A-c line filters at the source of interference, if the noise is created by an electrical appliance. (2) Noise -limiting circuits for the reduction, in the receiver itself, of interference of the type caused by automobile ignition systems. appliances, such as electric mixers, heating pads, vacuum sweepers, refrigerators, oil burners, sewing machines, doorbells, etc., create an interference of an intermittent nature. The insertion of a line filter near the source of interference often will effect a complete cure. Filters for small appliances can consist of a 0.1 -µfd. capacitor connected across the 110 -volt a -c line. Two capacitors in series across the line, with the midpoint connected to ground, can be used in conjunction with ultraviolet ray machines, refrigerators, oil burner furnaces, and other more stubborn offenders. In severe cases of interference, additional filters in the form of heavy duty r -f choke coils must be connected in series with the 110 -volt a -c line on both sides of the line right at the interfering appliance. Power Line Filters Many household Numerous noise -limiting circuits which are beneficial in overcoming key clicks, automobile ignition interference, and similar noise impulses have become popular. They operate on the Peak Noise Limiters principle that each individual noise pulse is 226 Radio of very short duration, yet of very high amplitude. The popping or clicking type of noise from electrical ignition systems may produce a signal having a peak value ten to twenty times as great as the incoming radio signal, but an average power much less than the signal. As the duration of this type of noise peak is short, the receiver can be made inoperative during the noise pulse without the human ear detecting the total loss of signal. Some noise limiters actually punch a bole in the signal, while others merely limit the maximum peak signal which reaches the headphones or loud- speaker. The noise peak is of such short duration that it would not be objectionable except for the fact that it produces an over -loading effect on the receiver, which increases its time constant. A sharp voltage peak will give a kick to the diaphragm of the headphones or speaker, and the momentum or inertia keeps the diaphragm in motion until the dampening of the diaphragm stops it. This movement produces a popping sound which may completely obliterate the desired signal. If the noise pulse can be limited to a peak amplitude equal to that of the desired signal, the resulting interference is practically negligible for moderately low repetition rates, such as ignition noise. In addition, the i -f amplifier of the receiver will also tend to lengthen the duration of the noise pulses because the relatively high -Q i -f tuned circuits will ring or oscillate when excited by a sharp pulse, such as produced by ignition noise. The most effective noise limiter would be placed before the high -Q i -f tuned circuits. At this point the noise pulse is the sharpest and has not been degraded by passage through the i -f transformers. In addition, the pulse is eliminated before it can produce ringing effects in the i -f chain. noise limiter is shown in figure 28. This is an adaptation of the Lamb noise silencer circuit. The i-f signal is fed into a double grid tube, such as a 6L7, and thence into the i -f chain. A 6AB7 high gain pentode is capacity coupled to the input of the i -f system. This auxiliary tube amplifies both signal and noise that is fed to it. It has a minimum of selectivity ahead of it so that it receives the true noise pulse before it is degraded by the i -f strip. A broadly tuned i -f transformer is used to couple the noise amplifier to a 6H6 noise rectifier. The gain of the noise amplifier is controlled by a potentiometer in the cathode of the 6AB7 noise amplifier. This potentiometer controls the gain of the noise amplifier The Lomb Noise Limiter THE Receiver Fundamentals An i -f IST DET RADIO ISTI.F. 2ND I.F. 617 Figure 28 THE LAMB I -F NOISE SILENCER stage and in addition sets the bias level on the 6H6 diode so that the incoming signal will not be rectified. Only noise peaks louder than the signal can overcome the resting bias of the 6H6 and cause it to conduct. A noise pulse rectified by the 6H6 is applied as a negative voltage to the control grid of the 6L7 i -f tube, disabling the tube, and punching a hole in the signal at the instant of the noise pulse. By varying the bias control of the noise limiter, the negative control voltage applied to the 6L7 may be adjusted until it is barely sufficient to overcome the noise impulses applied to the al control grid without allowing the modulation peaks of the carrier to become badly distorted. effective i -f noise limiter is the Bishop limiter. This is a full -wave shunt type diode limiter applied to the primary of the last i -f transformer of a receiver. The limiter is self- biased and automatically adjusts itself to the degree of modulation of the received signal. The schematic of this limiter is shown in figure 29. The bias circuit time constant is determined by C, and the shunt resistance, The Bishop Another Noise Limiter which consists of R, and R2 in series. The plate resistance of the last i -f tube and the capacity of C, determine the charging rate of the circuit. The limiter is disabled by opening which allows the bias to rise to the value of the i -f signal. S www.americanradiohistory.com HANDBOOK Noise Limiters 227 ter peak noise suppression than a standard communications receiver having an i -f bandwidth of perhaps 8 kc. Likewise, when a crystal filter is used on the "sharp" position an a -f peak' limiter is of little benefit. Practical Noise limiters range all the way from an audio stage Limiter Circuits running at very low screen or plate voltage, to elaborate affairs employing 5 or more tubes. Rather than attempt to show the numerous types, many of which are quite complex considering the results obtained, only two very similar types will be described. Either is just about as effective as the most elaborate limiter that can be constructed, yet requires the addition of but a single diode and a few resistors and capacitors over what would be employed in a good superheterodyne without a limiter. Both circuits, with but minor modifications in resistance and capacitance values, are incorporated in one form or another in different types of factory -built communications receivPeak Noise Figure 29 THE BISHOP I -F NOISE LIMITER Audio Noise Limiters of the simplest and most practical peak limiters for radioSome telephone reception employ one or two diodes either as shunt or series limiters in the audio system of the receiver. when a noise pulse exceeds a certain predetermined threshold value, the limiter diode acts either as a short or open circuit, depending upon whether it is used in a shunt or series circuit. The threshold is made to occur at a level high enough that it will not clip modulation peaks enough to impair voice intelligibility, but low enough to limit the noise peaks effectively. Because the action of the peak limiter is needed most on very weak signals, and these usually are not strong enough to produce proper a -v-c action, a threshold setting that is correct for a strong phone signal is not correct for optimum limiting on very weak signals. For this reason the threshold control often is tied in with the a -v -c system so as to make the optimum threshold adjustment automatic instead of manual. Suppression of impulse noise by means of an audio peak limiter is best accomplished at the very front end of the audio system, and for this reason the function of superheterodyne second detector and limiter often are combined in a composite circuit. The amount of limiting that can be obtained is a function of the audio distortion that can be tolerated. Because excessive distortion will reduce the intelligibility as much as will background noise, the degree of limiting for which the circuit is designed has to be a compromise. Peak noise limiters working at the second detector are much more effective when the i -f bandwidth of the receiver is broad, because a sharp i -f amplifier will lengthen the pulses by the time they reach the second detector, making the limiter less effective. V-h -f super heterodynes have an i -f bandwidth considerably wider than the minimum necessary for voice sidebands (to take care of drift and instability). Therefore, they are capable of bet- ers. Referring to figure 30, the first circuit shows conventional superheterodyne second detector, a.v.c., and first audio stage with the addition of one tube element, which may be either a separate diode or part of a twindiode as illustrated. Diode D, acts as a series gate, allowing audio to get to the grid of the a -f tube only so long as the diode is conducting. The diode is biased by a d -c voltage obtained in the same manner as a -v -c control voltage, the bias being such that pulses of short duration no longer conduct when the pulse voltage exceeds the carrier by approximately 60 per cent. This also clips voice modulation peaks, but not enough to impair intelligibility. It is apparent that the series diode clips only positive modulation peaks, by limiting upward modulation to about 60 per cent. Negative or downward peaks are limited automatically to 100 per cent in the detector, because obviously the rectified voltage out of the diode detector cannot be less than zero. Limiting the downward peaks to 60 per cent or so instead of 100 per cent would result in but little improvement in noise reduction, and the results do not justify the additional components required. It is important that the exact resistance values shown be used, for best results, and that 10 per cent tolerance resistors be used for R, and R,. Also, the rectified carrier voltage developed across C, should be at least 5 volts for good limiting. The limiter will work well on c -w telegraphy if the amplitude of beat frequency oscillator injection is not too high. Variable injection is to be preferred, adjustable from the front panel. a D 228 THE Receiver Fundamentals Radio V2 VI LAST I. F.TUDC I -0.1 -µfd. paper mice C_100 -44fd. mica C4, Cs -0.01 -µfd. paper megohm, Rr, R2 15 watt R,, R4- 220,000 ohms, Vt watt R6, R6-1 megohm, C1 AUDIO FT. RADIO C, -50 -4µtd. -1 4T Rr watt -2- megohm potentiometer Figure 30 NOISE LIMITER CIRCUIT, WITH ASSOCIATED A -V -C This limiter is of the series type, and is self-adjusting to carrier strength for phone reception. For proper operation several volts should be developed across the secondary of the last i -f transformer (IFT) under carrier conditions. If this feature is not provided, the b -f-o injection should be reduced to the lowest value that will give a satisfactory beat. When this is done, effective limiting and a good beat can be obtained by proper adjustment of the r -f and a -f gain controls. It is assumed, of course, that the a.v.c. is cut out of the circuit for c -w telegraphy reception. Alternative Limiter Circuit The more circuit of figure 31 is effective than that shown in figure 30 under certain conditions and requires the addition of only one more resistor and one more capacitor than the other circuit. Also, this circuit involves a smaller loss in output level than the circuit of figure 30. This circuit can be used with equal effectiveness with a combined diode - triode or diode -pentode tube (6R7, 6SR7, 6Q7, 6SQ7 or similar diode -triodes, or 6B8, 6SF7, or similar diode -pentodes) as diode detector and first audio stage. However, a separate diode must be used for the noise limiter, D,. This diode may be one -half of a 6H6, 6AL5, 7A6, etc., or it may be a triode connected 6J5, 6C4 or similar type. Note that the return for the volume control must be made to the cathode of the detector diode (and not to ground) when a dual tube is used as combined second- detector first- audio. This means that in the circuit shown in figure 31 a connection will exist across the points where the "X" is shown on the diagram since a common cathode lead is brought out of the tube for Dr and V,. If desired, of course, a single dual diode may be used for Dr and D, in this circuit as well as in the circuit of figure 30. Switching the limiter in and out with the switch S brings about no change in volume. In any diode limiter circuit such as the ones shown in these two figures it is important that the mid -point of the heater potential for the noise -limiter diode be as close to ground potential as possible. This means that the center -tap of the heater supply for the tubes should be grounded wherever possible rather than grounding one side of the heater supply as is often done. Difficulty with hum pickup in the limiter circuit may be encountered when one side of the heater is grounded due to the high values of resistance necessary in the limiter circuit. The circuit of figure 31 has been used with excellent success in several home -constructed receivers, and in the BC- 312/BC -342 and BC348 series of surplus communications receivers. It is also used in certain manufactured receivers. An excellent check on the operation of the noise limiter in any communications receiver can be obtained by listening to the Loran signals in the 160 -meter band. With the limiter out a sharp rasping buzz will be obtained when one of these stations is tuned in. With the noise limiter switched into the circuit the buzz should be greatly reduced and a low pitched hum should be heard. The most satisafctory diode noise limiter is the series full wave limiter, shown in figure 32. The positive noise peaks are clipped by diode A, the clipping level of which may be adjusted to clip at any modulation level between 25 per cent and 100 per cent. The negaative noise peaks are clipped by diode B at The Full -Wave Limiter a fixed level. Twin Noise Squelch. popularized by CQ magazine, is a combination of a diode noise clipper and an audio squelch tube. The squelch cit.- The TNS Limiter The HANDBOOK U -H -F Circuits 229 This circuit is of the selfadjusting type and gives less distortion for a given degree of modulation than the more limiter circuits. -470K, 14 won R3 -100K, watt common R1, R2 R., RS V2 -1 megohm, VI watt megohm potentiometer -2mica (approx.) c2- 0.01 -µtd. paper R6 C1- 0.00025 C3- 0.01 -4fd. C4- 0.01 -12fd. paper paper 02 -6116, D3, 6AL5, sections of diode 7A6, a or 6S8 -GT Figure 31 ALTERNATIVE NOISE LIMITER CIRCUIT cuit is useful in eliminating the grinding background noise that is the residual left by the diode clipper. In figure 33, the setting of the 470K potentiometer determines the operating level of the squelch action and should be set to eliminate the residual background noise. Because of the low inherent distortion of the TNS, it may be left in the circuit at all times. As with other limiters, the TNS requires a high signal level at the second detector for maximum limiting effect. 12 -10 Special Considerations wavelength sections of parallel conductors or concentric transmission line are not only more efficient but also become of practical dimensions. Tuning Short Lines Tubes and tuning capacitors con netted to the open end of a transmission line provide a capacitance that makes the resonant length less than a quarter wave -length. The amount of shortening for a specified capacitive reactance is determined by the surge impedance of the line in U -H -F Receiver Design increasingly higher frequencies, it becomes progressively more difficult to obtain a satisfactory amount of selectivity and impedance from an ordinary coil and capacitor used as a Transmission At F STAGE 2ND DEI. -AUDIO Line Circuits S n. DIO resonant circuit. On the other hand, quarter F T 2ND DET AuD10 Figure 32 THE FULL -WAVE SERIES AUDIO NOISE LIMITER Figure 33 THE TNS AUDIO NOISE LIMITER www.americanradiohistory.com 230 Radio Receiver Fundamentals THE RADIO CAVITY CAVITY 0LOOV LINE CONCENTRIC LINE CAVITY Figure 34 COUPLING AN ANTENNA TO A O © GVITY CAVITY GRIDS ELECTRON BEAM HOLE COAXIALRESONANT CIRCUIT (A) shows the recommended method for coupling a coaxial line to a coaxial resonant circuit. (B) shows an alternative method for use with an open wire type of antenna feed line. section. It is given by the equation for resonance: 1 2rr /C = Z. tan 1 77 = 3.1416, / is the frequency, C the capacitance, Z. the surge impedance of the line, and tan / is the tangent of the electrical length in degrees. The capacitive reactance of the capacitance across the end is 1 /(277 / C) ohms. For resonance, this must equal the surge impedance of the line times the tangent of its electrical length (in degrees, where 90° equals a quarter wave). It will be seen that twice the capacitance will resonate a line if its surge impedance is halved; also that a given capacitance has twice the loading effect when the frequency is doubled. in which It is possible to couple into a parallel -rod line by tapping directly on one or both rods, preferably through blocking capacitors if any d.c. is present. More commonly, however, a hairpin is inductively coupled at the shorting bar end, either to the bar or to the two rods, or both. This normally will result in a balanced load. Should a loop unbalanced to ground be coupled in, any resulting unbalance reflected into the rods can be reduced with a simple Faraday screen, made of a few parallel wires placed between the hairpin loop and the rods. These should be soldered at only one end and grounded. An unbalanced tap on a coaxial resonant Coupling Into Lines and Coaxial Circuits Figure 35 METHODS OF EXCITING A RESONANT CAVITY circuit can be made directly on the inner conductor at the point where it is properly matched (figure 34). For low impedances, such as a concentric line feeder, a small one -half turn loop can be inserted through a hole in the outer conductor of the coaxial circuit, being in effect a half of the hairpin type recommended for coupling balanced feeders to coaxial resonant lines. The size of the loop and closeness to the inner conductor determines the impedance matching and loading. Such loops coupled in near the shorting disc do not alter the tuning appreciably, if not overcoupled. cavity is a closed resonant made of metal. It is known also as a rhumbatron. The cavity, having both inductance and capacitance, supersedes coil -capacitor and capacitance- loaded transmission -line tuned circuits at extremely high frequencies where conventional L and C components, of even the most refined design, prove impractical because of the tiny electrical and physical dimensions they must have. Microwave cavities have high Q factors and are superior to conventional tuned circuits. They may be employed in the manner of an absorption wavemeter or as the tuned circuit in other r -f test instruments, and in microwave transmitters and receivers. Resonant cavities usually are closed on all sides and all of their walls are made of electrical conductor. However, in some forms, small openings are present for the purpose of excitation. Cavities have been produced in several shapes including the plain sphere, Resonant Cavities A chamber www.americanradiohistory.com HANDBOOK TUNING U dimpled sphere, sphere with reentrant cones of various sorts, cylinder, prism (including cube), ellipsoid, ellipsoid -hyperboloid, doughnut- shape, and various reentrant types. In appearance, they resemble in their simpler forms metal boxes or cans. The cavity actually is a linear circuit, but one which is superior to a conventional coaxial resonator in the s -h -f range. The cavity resonates in much the same manner as does a barrel or a closed room with reflecting walls. Because electromagnetic energy, and the associated electrostatic energy, oscillates to and fro inside them in one mode or another, resonant cavities resemble wave guides. The mode of operation in a cavity is affected by the manner in which micro-wave energy is injected. A cavity will resonate to a large number of frequencies, each being associated with a particular mode or standing -wave pattern. The lowest mode (lowest frequency of operation) of a cavity resonator normally is the one used. The resonant frequency of a cavity may be varied, if desired, by means of movable plungers or plugs, as shown in figure 36A, or a movable metal disc (see figure 36B). A cavity that is too small for a given wavelength will not oscillate. The resonant frequencies of simple spherical, cylindrical, and cubical cavities may be calculated simply for one particular mode. Wavelength and cavity dimensions (in centimeters) are related by the following simple resonance formulae: Butterfly 231 DISC Figure 36 TUNING METHODS FOR CYLINDRICAL RESONANT CAVITIES = 2.6 x = 2.83 x = 2.28x radius half of radius 1 side Unlike the cavity resonator, which in its conventional form is a device which can tune over a relatively narrow band, the butterfly circuit is a tunable resonator which permits coverage of a fairly Circuit Circuits MOVEABLE SLUGS For Cylinder A, " Cube Ar " Sphere A, -H -F Figure 37 THE BUTTERFLY RESONANT CIRCUIT Shown at (A) is the physical appearance of the butterfly circuit as used in the v -h -f and lower u-h-f ronge. (B) shows an electrical representation of the circuit. wide u -h -f band. The butterfly circuit is very similar to a conventional coil -variable capacitor combination, except that both inductance and capacitance are provided by what appears to be a variable capacitor alone. The Q of this device is somewhat less than that of a concentric -line tuned circuit but is entirely adequate for numerous applications. Figure 37A shows construction of a single butterfly section. The butterfly- shaped rotor, from which the device derives its name, turns in relation to the unconventional stator. The two groups of stator "fins" or sectors are in effect joined together by a semi -circular metal band, integral with the sectors, which provides the circuit inductance. When the rotor is set to fill the loop opening (the position in which it is shown in figure 37A), the circuit inductance and capacitance are reduced to minimum. When the rotor occupies the position indicated by the dotted lines, the inductance and capacitance are at maximum. The tuning range of practical butterfly circuits is in the ratio of 1.5:1 to 3.5:1. Direct circuit connections may be made to points A and B. If balanced operation is desired, either point C or D will provide the electrical mid-point. Coupling may be effected by means of a small singleturn loop placed near point E or F. The butterfly thus permits continuous variation of both capacitance and inductance, as indicated by the equivalent circuit in figure 37B, while at the same time eliminating all pigtails and wiping contacts. Several butterfly sections may be stacked in parallel in the same way that variable capacitors are built up. In stacking these sections, the effect of adding inductances in parallel is to lower the total circuit inductance, while the addition of stators and rotors raises the total capacitance, as well as the ratio of maximum to minimum capacitance. www.americanradiohistory.com 232 Radio Receiver Fundamentals Butterfly circuits have been applied specifically to oscillators for transmitters, superheterodyne receivers, and heterodyne frequency meters in the 100 - 1000 -Mc. frequency range. The types of resonant circuits described in the previous paragraphs have largely replaced conventional coil -capacitor circuits in the range above 100 Mc. Tuned short lines and butterfly circuits are used in the range from about 100 Mc. to perhaps 3500 Mc., and above about 3500 Mc. resonant cavities are used almost exclusively. The resonant cavity is also quite generally employed in the 2000 -Mc. to 3500 -Mc. range. In a properly designed receiver, thermal agitation in the first tuned circuit is amplified by subsequent tubes and predominates in the output. For good signal-to- set -noise ratio, therefore, one must strive for a high -gain low noise r -f stage. Hiss can be held down by giving careful attention to this point. A mixer has about 0.3 of the gain of an r -f tube of the same type; so it is advisable to precede a mixer by an efficient r-f stage. It is also of some value to have good r -f selectivity before the first detector in order to reduce noises produced by beating noise at one frequency against noise at another, to produce noise at the intermediate frequency in a superheterodyne. The frequency limit of a tube is reached when the shortest possible external connections are used as the tuned circuit, except for abnormal types of oscillation. Wires or sizeable components are often best considered as sections of transmission lines rather than as simple resistances, capacitances, or inductReceiver Circuits ances. as small triodes and pentodes will operate normally, they are generally preferred as v-h -f tubes over other receiving methods that have been devised. However, the input capacitance, input conductance, and transit time of these tubes limit the upper frequency at which they may be operated. The input resistance, which drops to a low value at very short wave -lengths, limits the stage gain and broadens the tuning. So long The first tube in a v -h -f receiver is most important in raising the signal above the noise generated in successive stages, for which reason small v -h-f types are definitely preferred. Tubes employing the conventional grid-controlled and diode rectifier principles have been modernized, through various expedients, for operation at frequencies as high, in some new types, as 4000 Mc. Beyond that frequency, electron transit time becomes the limiting facV -H -F Tubes THE RADIO tor and new principles must be enlisted. In general, the improvements embodied in existing tubes have consisted of (1) reducing electrode spacing to cut down electron transit time, (2) reducing electrode areas to decrease interelectrode capacitances, and (3) shortening of electrode leads either by mounting the electrode assembly close to the tube base or by bringing the leads out directly through the glass envelope at nearby points. Through reduction of lead inductance and interelectrode capacitances, input and output resonant frequencies due to tube construction have bee': increased substantially. Tubes embracing one or more of the features just outlined include the later !octal types, high- frequency acorns, button -base types, and the lighthouse types. Type 6J4 button-base triode will reach 500 Mc. Type 6F4 acorn triode is recommended for use up to 1200 Mc. Type 1A3 button -base diode has a resonant frequency of 1000 Mc., while type 9005 acorn diode resonates at 1500 Mc. Lighthouse type 2C40 can be used at frequencies up to 3500 Mc. as an oscillator. More than two de c a d e s have passed since the crystal (mineral) rectifier enjoyed widespread use in radio receivers. Low -priced tubes completely supplanted the fragile and relatively insensitive crystal detector, although it did continue for a few years as a simple meter rectifier in absorption wavemeters after its demise as a receiver component. Today, the crystal detector is of new importance in microwave communication. It is being employed as a detector and as a mixer in receivers and test instruments used at ex- Crystal Rectifiers tremely high radio frequencies. At some of the frequencies employed in microwave operations, the crystal rectifier is the only satisfactory detector or mixer. The chief advantages of the crystal rectifier are very low capacitance, relative freedom from transit -time difficulties, and its two -terminal nature. No batteries or a -c power supply are required for its operation. The crystal detector consists essentially of a small piece of silicon or germanium mounted in a base of low- melting -point alloy and contacted by means of a thin, springy feeler wire known as the cat whisker. This arrangement is shown in figure 38A. The complex physics of crystal rectification is beyond the scope of this discussion. It is sufficient to state that current flows from several hundred to several thousand times more readily in one direction through the contact of cat whisker and crystal than in the opposite direction. Consequently, an alternating current (including one of microwave frequency) will www.americanradiohistory.com HANDBOOK SYMBOL f Receiver Adjustment \'i BRASS BASE CONNECTOR -CERAMIC SLEEVE BRASS CAP BRASS CONNECTOR PIN CRYSTAL DIODE small silicon crystal is attached to the base connector and o fine "cat whisker" wire is set to the most sensitive spot on the crystal. After adjustment the ceramic shell is filled with compound to hold the contact wire in position. Crystals of this type are used to over 30,000 mc. A rectified by the crystal detector. The load, through which the rectified currents flow, may be connected in series or shunt with the crystal, although the former connection is most generally employed. The basic arrangement of a modern fixed crystal detector developed during World War II for microwave work, particularly radar, is shown in figure 38B. Once the cat whisker of this unit is set at the factory to the most sensitive spot on the surface of the silicon crystal and its pressure is adjusted, a filler compound is injected through the filling hole to hold the cat whisker permanently in position. be Receiver Adjustment simple regenerative receiver requires adjustment other than that necessary to insure correct tuning and smooth regeneration over some desired range. Receivers of the tuned radio- frequency type and superheterodynes require precise alignment to obtain the highest possible degree of selectivity and A little sensitivity. Good results can be obtained from a receiver only when it is properly aligned and adjusted. The most practical technique for mak- ing these adjustments is given below. A very small number ments Alignment procedure in a multistage t -r -f receiver is exactly the same as aligning a single stage. If the detector is regenerative, each preceding stage is successively aligned while keeping the detector circuit tuned to the test signal, the latter being a station signal or one locally generated by a test oscillator loosely coupled to the antenna lead. During these adjustments, the r -f amplifier gain control is adjusted for maximum sensitivity, assuming that the r-f amplifier is stable and does not oscillate. Often a sensitive receiver can be roughly aligned by tuning for maximum noise pickup. Alignment 1N23 MICROWAVE -TYPE Instruments receiver output when tuning to a modulated signal. If the signal is a steady tone, such as from a test oscillator, the output meter will indicate the value of the detected signal. In this manner, alignment results may be visually noted on the meter. T -R -F Receiver Figure 38 12 -11 233 of instru- will suffice to check and align a communications receiver, the most important of these testing units being a modulated oscillator and a d -c and a -c voltmeter. The meters are essential in checking the voltage applied at each circuit point from the power supply. If the a -c voltmeter is of the oxide rectifier type, it can be used, in addition, as an output meter when connected across the Aligning a superhet is a detailed task requiring a great amount of care and patience. It should never be undertaken without a thorough understanding of the involved job to be done and then only when there is abundant Superheterodyne Alignment time to devote to the operation. There are no short cuts; every circuit must be adjusted individually and accurately if the receiver is to give peak performance. The precision of each adjustment is dependent upon the accuracy with which the preceding one was made. Superhet alignment requires (1) a good signal generator (modulated oscillator) covering the radio and intermediate frequencies and equipped with an attenuator; (2) the necessary socket wrenches, screwdrivers, or "neutralizing tools" to adjust the various i -f and r -f trimmer capacitors; and (3) some convenient type of tuning indicator, such as a copper oxide or electronic voltmeter. Throughout the alignment process, unless specifically stated otherwise, the r -f gain control must be set for maximum output, the beat oscillator switched off, and the a.v.c. turned off or shorted out. When the signal output of the receiver is excessive, either the attenuator or the a -f gain control may be turned down, but never the r -f gain control. After the receiver has been given a rigid electrical and mechanical inspection, and any faults which may have been found in wiring or the selection and assembly of parts corrected, the i -f amplifier may be aligned as the first step in the checking operations. Vl ith the signal generator set to give a modulated signal on the frequency at which the i -f I -F Alignment www.americanradiohistory.com 234 amplifier is to operate, clip the "hot" output lead from the generator to the last i -f stage through a small fixed capacitor to the control grid. Adjust both trimmer capacitor's in the last i -f transformer (the one between the last i -f amplifier and the second detector) to resonance as indicated by maximum deflection of the output meter. Each i -f stage is adjusted in the same manner, moving the hot lead, stage by stage, back toward the front end of the receiver and backing off the attenuator as the signal strength increases in each new position. The last adjustment will be made to the first i -f transformer, with the hot signal generator lead connected to the control grid of the mixer. Occasionally it is necessary to disconnect the mixer grid lead from the coil, grounding it through a 1,000- or 5,000 -ohm resistor, and coupling the signal generator through a small capacitor to the grid. When the last i -f adjustment has been completed, it is good practice to go back through the i -f channel, re- peaking all of the transformers. It is imperative that this recheck be made in sets which do not include a crystal filter, and where the simple alignment of the i -f amplifier to the generator is final. I RADIO THE Radio Receiver Fundamentals -F SIGNAL IN ALONE t w ` F PLUS O MULTIPLIER f W f 55 CC FREQUENCY Figure 39 THE Q- MULTIPLIER The loss resistance of a high -Q neutralized by regeneration is circuit in a feedback amplifier. A highly selective passband is produced which circuit of the is coupled to the i -f receiver. simple -F with Crystal Filter There are several ways of align ing an i -f channel which contains a crystal -filter circuit. However, the following method is one which has been found to give satisfactory results in every case: An unmodulated signal generator capable of tuning to the frequency of the filter crystal in the receiver is coupled to the grid of the stage which precedes the crystal filter in the receiver. Then, with the crystal filter switched in, the signal generator is tuned slowly to find the frequency where the crystal peaks. The receiver "S" meter may be used as the indicator, and the sound heard from the loudspeaker will be of assistance in finding the point. When the frequency at which the crystal peaks has been found, all the i -f transformers in the receiver should be touched up to peak at that frequency. I Adjusting the beat oscillator on a receiver that has no front panel adjustment is relatively simple. It is only necessary to tune the receiver to resonance with any signal, as indicated by the tuning indicator, and then turn on the b.f.o. B -F -Q Adjustment and set its trimmer (or trimmers) to produce the desired beat note. Setting the beat oscil- lator in this way will result in the beat note being stronger on one "side" of the signal than on the other, which is what is desired for c -w reception. The b.f.o. should not be set to zero beat when the receiver is tuned to resonance with the signal, as this will cause an equally strong beat to be obtained on both sides of resonance. Alignment of the front end of a receiver is a relatively simple process, consisting of first getting the oscillator to cover the desired frequency range and then of peaking the various r -f circuits for maximum gain. However, if the frequency range covered by the receiver is very wide a fair amount of cut and try will be required to obtain satisfactory tracking between the r-f circuits and the oscillator. Manufactured communications receivers should always be tuned in accordance with the instructions given in the maintenance manual for the receiver. Front -End Alignment 12 -12 home -constructed Receiving Accessories The selectivity of a receiver may be increased by raising the Q of the tuned circuits of the i -f strip. A simple way to accomplish this is to add a controlled amount of positive feedback to a tuned circuit, thus increasing its Q. This is done in the 0- multiplier, whose basic circuit is shown in figure 39. The circuit L -C1 -C2 is tuned to The Q- Multiplier www.americanradiohistory.com Receiving Accessories 235 HANDßCOK I.F SIGNAL IN I -F SIGNAL OJT .005 6 _ L2 Lt .005 NULL O 12 10X7 TO PLATE TERMINAL OF FIRST I-F TUBE TNRU 2. OF COAXIAL LINE PEAR MULTIPLIER "NULL" SELECTIfIrY 2 12Ax7 CONTROLS 3 32 MEG 1.SK 10K I LI. GRAY6'URNE 455 KC FREQUENCY Figure MULTIPLIER NULL CIRCUIT 1- n T 6.3 V. (0.6-6.OAIN) intermediate frequency, and the loss resistance of the circuit is neutralized by the positive feedback circuit composed of C3 and the vacuum tube. Too great a degree of positive feedback will cause the circuit to break into LI is required to tune out the reactance of the coaxial line. It is adjusted for maximum signal response. LI may be omitted if the Q- multiplier is connected to the receiver with o short length of wire, and the i -f transformer within the receiver is retuned. Coil the oscillation. At the resonant frequency, the impedance of the tuned circuit is very high, and when shunted across an i -f stage will have little effect upon the signal. At frequencies removed from resonance, the impedance of the circuit is low, resulting in high attenuation of the i -f signal. The resonant frequency of the Q- multiplier may by varied by changing the value of one of the components in the tuned circuit. The Q- multiplier may also be used to "null" a signal by employing negative feedback to control the plate resistance of an auxiliary amplifier stage as shown in figure 40. Since the grid- cathode phase shift through the Q- multiplier is zero, the plate resistance of a second tube may be readily controlled by placing it across the Q- multiplier. At resonance, the high negative feedback drops the plate resistance of V2, shunting the i -f circuit. Off resonance, the feedback is reduced and the plate resistance of V2 rises, reducing the amount of signal attenuation in the i -f strip. A circuit combining both the "peak" and "null" features is shown in figure 41. A version of the common mixer or converter stage 41 SCHEMATIC OF A 455KC 0- MULTIPLIER The addition of a second triode permits the 0- Multiplier to be used for nulling out an unwanted hetrodyne. The Product Detector V. L22GRAY6URNE "LOOPSTICK" COIL Figure 40 Q- +200-300 B V6 CNOKE A. may be used as a second detector in a receiver in place of the usual diode detector. The diode is an envelope detector (section 12 -1) and develops a d -c output voltage from a single r -f signal, and audio "beats" from two or more input signals. A product detector (figure 42) requires that a local carrier voltage be present in order to produce an audio output signal. R PRODUCT DETECTOR -F SIGNAL LOCAL OSCILLATOR Figure 42 THE PRODUCT DETECTOR Audio output signal is when developed only local oscillator is on. www.americanradiohistory.com HAUDIO OUTPUT Radio Receiver Fundamentals 236 Va V, I -F SIGNAL VS 12AU7 12AU7 I-F 0, r+AU01O OUT 4711 SIC. SO SEAT OSC. SIGNAL Figure Figure 43 PENTAGRID MIXER USED AS PRODUCT DETECTOR PRODUCT DETECTOR VI and V2 act as cathode followers, delivering sideband signal and local oscillator signal to grounded grid triode mixer (V3). Such a detector is useful for single sideband work, since the inter -modulation distortion is extremely low. A pentagrid product detector is shown in figure 43. The incoming signal is applied to grid 3 of the mixer tube, and the local oscillator is injected on grid 1. Grid bias is adjusted for operation over the linear portion of the tube characteristic curve. When grid 1 injection is removed, the audio output from an unmodulated signal applied to grid 3 should be reduced approximately 30 to 40 db below normal detection level. When the frequency of the local oscillator is synchronized with the incoming carrier, amplitude modulated signals may be received by exalted carrier reception, wherein the local carrier substitutes for the transmitted carrier of the a -m signal. Three triodes may be used as a product detector (figure 44). Triodes V1 and V2 act as cathode followers, delivering the sideband signal and the local oscillator signal to a grounded grid triode (V3) which functions as the mixer stage. A third version of the product detector is illustrated in figure 45. A twin triode tube is used. Section V1 functions as a cathode follower amplifier. Section V2 is a . ?tl ì II14611 i.h°1 .. IIIIII 44 TRIPLE -TRIODE Figure 45 DOUBLE -TRIODE PRODUCT DETECTOR "plate" detector, the cathode of which is with the cathode follower amplifier. The local oscillator signal is injected into the grid circuit of tube V2. common Figure 46 EXPLODED VIEW OF COLLINS MECHANICAL FILTER .1. 'If#I!IIItIYIIiIIIIvIIIII'I+II vIIl°1i www.americanradiohistory.com 1 1 CHAPTER THIRTEEN Generation of Radio Frequency Energy A radio communication or broadcast transmitter consists of a source of radio frequency power, or carrier; a system for modulating the carrier whereby voice or telegraph keying or other modulation is superimposed upon it; and an antenna system, including feed line, for radiating the intelligence- carrying radio frequency power. The power supply employed to convert primary power to the various voltages required by the r -f and modulator portions of the transmitter may also be considered part of the transmitter. Voice modulation usually is accomplished by varying either the amplitude or the frequency of the radio frequency carrier in accordance with the components of intelligence to be transmitted. Radiotelegraph modulation (keying) normally is accomplished either by interrupting, shifting the frequency of, or superimposing an audio tone on the radio -frequency carrier in accordance with the dots and dashes to be transmitted. The complexity of the radio- frequency generating portion of the transmitter is dependent upon the power, order of stability, and frequency desired. An oscillator feeding an antenna directly is the simplest form of radio- frequency generator. A modern high -frequency transmitter, on the other hand, is a very complex generator. Such an equipment usually comprises a very stable crystal -controlled or self-controlled oscillator to stabilize the output frequency, a series of frequency multipliers, one or more amplifier stages to increase the power up to the level which is desired for feeding the antenna system, and a filter system for keeping the harmonic energy generated in the transmitter from being fed to the antenna system. 13 -1 Controlled Oscillators Self- In Chapter Four, it was explained that the amplifying properties of a tube having three or more elements give it the ability to generate an alternating current of a frequency determined by the components associated with it. A vacuum tube operated in such a circuit is called an oscillator, and its function is essentially to convert direct current into radio frequency alternating current of a predetermined frequency. Oscillators for controlling the frequency of conventional radio transmitters can be divided into two general classes: self-controlled and crystal- controlled. There are a great many types of self-controlled oscillators, each of which is best suited 237 www.americanradiohistory.com Generation of 238 OA SHUNT -FED HARTLEY R -F Energy THE OB SHUNT -FED COLPITTS R © RADIO TUNED PLATE TUNED GRID R L+ 250 Li 250 GRID COIL OD TUNED -PLATE UNTUNED GRID EO ELECTRON COUPLED HO CLAPP ELECTRON COUPLED FO COLPITTS ELECTRON COUPLED L OG CLAPP Figure 1 COMMON TYPES OF SELF -EXCITED OSCILLATORS Fixed capacitor values are typical, but will vary somewhat with the application. In the Clapp oscillator circuits (G) and (H), capacitors Cr and C2 should have reactance of 50 to 100 ohms at the operating frequency of the oscillator. Tuning ofa these two oscillators is accomplished by capacitor C. In the circuits of (E), (F), and (H), tuning of the tank circuit in the plate of the oscillator tube will have relatively small effect on the frequency of oscillation. The plate tank circuit also may, if desired, be tuned to a harmonic of the oscillation frequency, or a broadly resonant circuit may be used in this circuit position. to a particular application. They can further be subdivided into the classifications of: neg- ative -grid oscillators, electron -orbit oscillators, negative -resistance oscillators, velocity modulation oscillators, and magnetron oscillators. negative -grid oscillator is a vacuum-tube amplifier with a sufficient portion of the output energy coupled back into the input circuit to sustain oscillation. The conNegative -Grid A Oscillators essentially trol grid is biased negatively with respect to the cathode. Common types of negative -grid oscillators are diagrammed in figure 1. Illustrated in figure 1 (A) is the oscillator circuit which finds the most general application at the present time; this circuit is commonly called the Hartley. The operation of this oscillator will be described as an index to the operation of all negative -grid oscillators; the only real differThe Hartley www.americanradiohistory.com Oscillators HANDBOOK ence between the various circuits is the manner in which energy for excitation is coupled from the plate to the grid circuit. When plate voltage is applied to the Hartley oscillator shown at (A), the sudden flow of plate current accompanying the application of plate voltage will cause an electro- magnetic field to be set up in the vicinity of the coil. The building -up of this field will cause a potential drdp to appear from turn-to -turn along the coil. Due to the inductive coupling between the portion of the coil in which the plate current is flowing and the grid portion, a potential will be induced in the grid portion. Since the cathode tap is between the grid and plate ends of the coil, the induced grid voltage acts in such a manner as to increase further the plate current to the tube. This action will continue for a short period of time determined by the inductance and capacitance of the tuned circuit, until the flywheel effect of the tuned circuit causes this action to come to a maximum and then to reverse itself. The plate current then decreases, the magnetic field around the coil also decreasing, until a minimum is reached, when the action starts again in the original direction and at a greater amplitude than before. The amplitude of these oscillations, the frequency of which is determined by the coil -capacitor circuit, will increase in a very short period of time to a limit determined by the plate voltage of the oscil- lator tube. (8) shows a version of oscillator. It can be seen that this is essentially the same circuit as the Hartley except that the ratio of a The Colpitts Figure 1 the Colpitts pair of capacitances in series determines the effective cathode tap, instead of actually using a tap on the tank coil. Also, the net capacitance of these two capacitors comprises the tank capacitance of the tuned circuit. This oscillator circuit is somewhat less susceptible to parasitic (spurious) oscillations than the Hartley. For best operation of the Hartley and Colpitts oscillators, the voltage from grid to cathode, determined by the tap on the coil or the setting of the two capacitors, normally should be from 1/3 to 1/5 that appearing between plate and cathode. The tuned -plate tuned-grid oscillator illustrated at (C) has a tank circuit in both the plate and grid circuits. The feedback of energy from the plate to the grid circuits is accomplished by the The T.P.T.G. plate -to-grid inter -electrode capacitance within the tube. The necessary phase reversal in feedback voltage is provided by tuning the grid tank capacitor to the low side of the de- 239 sired frequency and the plate capacitor to the high side. A broadly resonant coil may be substituted for the grid tank to form the T.N. T. oscillator shown at (D). Electron -Coupled In any of the oscillator circuits just described it is possible to take energy from the oscillator circuit by coupling an external load to the tank circuit. Since the tank circuit determines the frequency of oscillation of the tube, any variations in the conditions of the external circuit will be coupled back into the frequency determining portion of the oscillator. These variations will result in frequency instability. The frequency determining portion of an oscillator may be coupled to the load circuit only by an electron stream, as illustrated in (E) and (F) of figure 1. When it is considered that the screen of the tube acts as the plate to the oscillator circuit, the plate merely actOscillators ing as a coupler to the load, then the similarity between the cathode -grid- screen circuit of these oscillators and the cathode -grid -plate circuits of the corresponding prototype can be seen. The electron- coupled oscillator has good stability with respect to load and voltage variation. Load variations have a relatively small effect on the frequency, since the only coupling between the oscillating circuit and the load is through the electron stream flowing through the other elements to the plate. The plate is electrostatically shielded from the oscillating portion by the bypassed screen. The stability of the e.c.o. with respect to variations in supply voltages is explained as follows: The frequency will shift in one direction with an increase in screen voltage, while an increase in plate voltage will cause it to shift in the other direction. By a proper proportioning of the resistors that comprise the voltage divider supplying screen voltage, it is possible to make the frequency of the oscillator substantially independent of supply voltage variations. The Clapp Oscillator relatively new type of oscillator circuit which is capable of giving excellent frequency stability is A illustrated in figure 1G. Comparison between the more standard circuits of figure IA through IF and the Clapp oscillator circuits of figures 1G and 1H will immediately show one marked difference: the tuned circuit which controls the operating frequency in the Clapp oscillator is series resonant, while in all the more standard oscillator circuits the frequency controlling circuit is parallel resonant. Also, the capacitors C, and C, are relatively large in terms of the usual values for a Colpitts oscil- 240 Generation of R -F THE Energy lator. In fact, the value of capacitors C, and C, will be in the vicinity of 0.001 µfd. to 0.0025 µfd. for an oscillator which is to be operated in the 1.8 -Mc. band. The Clapp oscillator operates in the following manner: at the resonant frequency of the oscillator tuned circuit (L, C) the impedance of this circuit is at minimum (since it operates in series resonance) and maximum current flows through it. Note however, that C, and C, also are included within the current path for the series resonant circuit, so that at the frequency of resonance an appreciable voltage drop appears across these capacitors. The voltage drop appearing across C, is applied to the grid of the oscillator tube as excitation, while the amplified output of the oscillator tube appears across C, as the driving power to keep the circuit in oscillation. Capacitors C, and C, should be made as large in value as possible, while still permitting the circuit to oscillate over the full tuning range of C. The larger these capacitors are made, the smaller will be the coupling between the oscillating circuit and the tube, and consequently the better will be oscillator stability with respect to tube variations. High Gm tubes such as the 6AC7, 6ÁG7, and 6CB6 will permit the use of larger values of capacitance at C, and C, than will more conventional tubes such as the 6SJ7, 6V6, and such types. In general it may be said that the reactance of capacitors C, and C, should be on the order of 40 to 120 ohms at the operating frequency of the oscillator -with the lower values of reactance going with high -Gm tubes and the higher values being necessary to permit oscillation with tubes having Gm in the range of 2000 micromhos such as the 6SJ7. It will be found that the Clapp oscillator will have a tendency to vary in power output over the frequency range of tuning capacitor C. The output will be greatest where C is at its largest setting, and will tend to fall off with C at minimum capacitance. In fact, if capacitors C, and C, have too large a value the circuit will stop oscillation near the minimum capacitance setting of C. Hence it will be necessary to use a slightly smaller value of capacitance at C, and C, (to provide an increase in the capacitive reactance at this point), or else the frequency range of the oscillator must be restricted by paralleling a fixed capacitor across C so that its effective capacitance at minimum setting will be increased to a value which will sustain oscillation. In the triode Clapp oscillator, such as shown at figure 1G, output voltage for excitation of an amplifier, doubler, or isolation stage normally is taken from the cathode of the oscillator tube by capacitive coupling to the grid of the next tube. However, where greater iso- RADIO lation of succeeding stages from the oscillating circuit is desired, the electron- coupled Clapp oscillator diagrammed in figure 1H may be used. Output then may be taken from the plate circuit of the tube by capacitive coupling with either a tuned circuit, as shown, or with an r -f choke or a broadly resonant circuit in the plate return. Alternatively, energy may be coupled from the output circuit L,-C, by link coupling. The considerations with regard to and the grid tuned circuit are the same as for the triode oscillator arrangement of CC figure 1G. Negative- resistance oscillators often are used when unusually high frequency stability is desired, as in a frequency meter. The dynatron of a few years ago and the newer transitron are examples of oscillator circuits which make use of the negative resistance characteristic between different elements in some multi -grid tubes. In the dynatron, the negative resistance is a consequence of secondary emission of electrons from the plate of a tetrode tube. By a proper proportioning of the electrode voltage, an increase in screen voltage will cause a decrease in screen current, since the increased screen voltage will cause the screen to attract a larger number of the secondary electrons emitted by the plate. Since the net screen current flowing from the screen supply will be decreased by an increase in screen voltage, it is said that the screen circuit presents a negative resistance. If any type of tuned circuit, or even a resistance- capacitance circuit, is connected in series with the screen, the arrangement will oscillate -provided, of course, that the external circuit impedance is greater than the negative resistance. A negative resistance effect similar to the dynatron is obtained in the transitron circuit, which uses a pentode with the suppressor coupled to the screen. The negative resistance in this case is obtained from a combination of secondary emission and inter -electrode coupling, and is considerably more stable than that obtained from uncontrolled secondary emission alone in the dynatron. A representative transitron oscillator circuit is shown in figure 2. The chief distinction between a conventional negative grid oscillator and a negative resistance oscillator is that in the former the tank circuit must act as a phase inverter in order to permit the amplification of the tube to act as a negative resistance, while in the latter the tube acts as its own phase inverter. Thus a negative resistance oscillator requires only an untapped coil and a single capacitor Negative Resistance Oscillators www.americanradiohistory.com HANDBOOK Oscillators 6SK7 TWO -TERMINAL 241 Figure 2 OSCILLATOR CIRCUITS Both circuits may be used for an audio oscillator or for frequencies into the v -h -f range simply by placing a tank circuit tuned to the proper frequency where indicated on the drawing. Recommended values for the components are given below for both oscillators. O 6SN7 TRANSITRON OSCILLATOR OR 6J6 TRANSITION OSCILLATOR 0.01 -µfd. mica for r.f. 10 -µfd. elect. for c.f. C2- 0.00005 -µfd. mica for r.f. 0.1 -4fd. paper for a.f. C3- 0.003-µfd. mico for r.f. 03-4fd. paper for a.f. C4- 001 -µfd. mica for r.f. 8 -4fd. elect. for a.f. R3-220K S5 -watt carbon R2 1800 ohms 55-watt carbon R3 -22K 2 -watt carbon R4 -22K 2 -watt carbon C,- CATHODE -COUPLED OSCILLATOR C3- 0.00005 -µfd. mica for r.f. 0.1 -µfd. paper for audio C2- 0.003 -µfd. mica for audio O R1-47K CATHODE COUPLED OSCILLATOR as the frequency determining tank circuit, and is classed as a two terminal oscillator. In fact, the time constant of an R/C circuit may be used as the frequency determining element and such an oscillator is rather widely used as a tunable audio frequency oscillator. The Franklin oscillator makes use of two cascaded tubes to obtain the negative- resistance effect (figure 3). The tubes may be either a pair of triodes, tetrodes, or pentodes, a dual The Fronklin Oscillator triode, or a combination of a triode and a multi grid tube. The chief advantage of this oscillator circuit is that the frequency determining tank only has two terminals, and one side of the circuit is grounded. The second tube acts as a phase inverter to give an effect similar to that obtained with the dynatron or transitron, except that the effective transconductance is much higher. If the tuned circuit is omitted or is replaced by a resistor, the circuit becomes a relaxation oscillator or a multivibrator. The Clapp oscillator has proved to be inherently the most stable of all the oscillator circuits discussed above, since minimum coupling between the oscillator tube and its associated tuned circuit is possible. However, this inOscillator Stability R2 -1K rd. 8 -pfd. lect. for -watt carbon -watt carbon S5 1 herently good stability is with respect to tube variations; instability of the tuned circuit with respect to vibration or temperature will of course have as much effect on the frequency of oscillation as with any other type of oscillator circuit. Solid mechanical construction of the components of the oscillating circuit, along with a small negative -coefficient compensating capacitor included as an element of the tuned circuit, usually will afford an adequate degree of oscillator stability. Figure 3 THE FRANKLIN OSCILLATOR CIRCUIT A separate phase inverter tube is used in this oscillator to feed a portion of the output back to the input in the proper phase to sustain oscillation. The values of Cr and C2 should be as small as will permit oscillations to be sustained over the desired frequency range. 242 Generation of R -F used to control the frequency of a transmitter in which there are stringent limitations on frequency tolerance, several precautions are taken to ensure that a variable frequency oscillator will stay on frequency. The oscillator is fed from a voltage regulated power supply, uses a well designed and temperature compensated tank circuit, is of rugged mechanical construction to avoid the effects of shock and vibration, is protected against excessive changes in ambient room temperature, and is isolated from feedback or stray coupling from other portions of the transmitter by shielding, filtering of voltage supply leads, and incorporation of one or more buffer- amplifier stages. In a high power transmitter a small amount of stray coupling from the final amplifier to the oscillator can produce appreciable degradation of the oscillator stability if both are on the same frequency. Therefore, the oscillator usually is operated on a subharmonic of the transmitter output frequency, with one or more frequency multipliers between the oscillator and final amplifier. V. F.O. Transmit - When ter Controls 13 -2 Quartz Crystal Oscillators Quartz is a naturally occuring crystal having a structure such that when plates are cut relationships to the crystallographic axes, these plates will show the piezoelectric effect -the plates will be deformed in the influence of an electric field, and, conversely, when such a plate is compressed or deformed in any way a potential difference will appear upon its opposite sides. The crystal has mechanical resonance, and will vibrate at a very high frequency because of its stiffness, the natural period of vibration depending upon the dimensions, the method of electrical excitation, and crystallographic orientation. Because of the piezoelectric properties, it is possible to cut a quartz plate which, when provided with suitable electrodes, will have the characteristics of a series resonant circuit with a very high L/C ratio and very high Q. The Q is several times as high as can be obtained with an inductor -capacitor combination in conventional physical sizes. The equivalent electrical circuit is shown in figure 4A, the resistance component simply being an acknowledgment of the fact that the Q, while high, does not have an infinite value. The shunt capacitance of the electrodes and associated wiring (crystal holder and socket, plus circuit wiring) is represented by the dotted portion of figure 4B. In a high frequency in certain definite THE Energy CI RADIO L (SMALL) Lt (LARGE) 4: C2 I (sraAr SHUNT) RI (SMALL) -J Figure 4 EQUIVALENT ELECTRICAL CIRCUIT OF QUARTZ PLATE IN A HOLDER At (A) is shown the equivalent series -resonant circuit of the crystal itself, at (B) is shown how the shunt capacitance of the holder electrodes and associated wiring affects the circuit to the combination circuit of (C) which exhibits both series resonance and parallel resonance (anti -resonance), the separation in frequency between the two modes being very small and determined by the ratio of C1 to C,. crystal this will be considerably greater than the capacitance component of an equivalent series L/C circuit, and unless the shunt capacitance is balanced out in a bridge circuit, the crystal will exhibit both resonant (series resonant) and anti- resonant (parallel resonant) frequencies, the latter being slightly higher than the series resonant frequency and ap- proaching it as C, is increased. The series resonance characteristic is employed in crystal filter circuits in receivers and also in certain oscillator circuits wherein the crystal is used as a selective feedback element in such a manner that the phase of the feedback is correct and the amplitude adequate only at or very close to the series resonant frequency of the crystal. While quartz, tourmaline, Rochelle salts, ADP, and EDT crystals all exhibit the piezoelectric effect, quartz is the material widely employed for frequency control. As the cutting and grinding of quartz plates has progressed to a high state of development and these plates may be purchased at prices which discourage the cutting and grinding by simple hand methods for one's own use, the procedure will be only lightly touched upon here. The crystal blank is cut from the raw quartz at a predetermined orientation with respect to the optical and electrical axes, the orientation determining the activity, temperature coefficient, thickness coefficient, and other characteristics. Various orientations or "cuts" having useful characteristics are illustrated in figure 5. www.americanradiohistory.com Crystal Oscillators HANDBOOK neo 243 T[.neror[ co rmus..T eD fllT[eS fe[0U[CT A7.11 OSCILLATORS MK.. LOW feCWCMCT AT ST Figure 5 CT 34. ¡ .N` T CT.DT,[T,fT SSiS' IT -ST. L er - Ta b 2[1.10 Se URUC `«e. s.ocs , ORIENTATION OF THE y COMMON CRYSTAL CUTS Tóir[eTUet Xff: ó veD.tl ^p.¢ xo.o 27e0 a` OSCILLATORS L 1 D -' > 1 ' a r ZERO o nA OXillTOe3 .r.D filTCeS p.c...., cDC The crystal blank is then rough -ground almost to frequency, the frequency increasing in inverse ratio to the oscillating dimension (usually the thickness). It is then finished to exact frequency either by careful lapping, by etching, or plating. The latter process consists of finishing it to a frequency slightly higher than that desired and then silver plating the electrodes right on the crystal, the frequency decreasing as the deposit of silver is increased. If the crystal is not etched, it must be carefully scrubbed and "baked" several times to stabilize it, or otherwise the frequency and activity of the crystal will change with time. Irradiation by X -rays recently has been used in crystal finishing. Unplated crystals usually are mounted in pressure holders, in which two electrodes are held against the crystal faces under slight pressure. Unplated crystals also are sometimes mounted in an air -gap holder, in which there is a very small gap between the crystal and one or both electrodes. By making this gap variable, the frequency of the crystal may be altered over narrow limits (about 0.3% for certain types). The temperature coefficient of frequency for various crystal cuts of the " -T" rotated family is indicated in figure 5. These angles are typical, but crystals of a certain cut will vary slightly. By controlling the orientation and dimensioning, the turning point (point of zero temperature coefficient) for a BT cut plate may be made either lower or higher than the 75 degrees shown. Also, by careful control of axes and dimensions, it is possible to get AT cut crystals with a very flat temperature- frequency characteristic. S, CCnTUT .T f ..[ee , . _ lTtes rr - [. The first quartz plates used were either Y cut or X cut. The former had a very high temperature coefficient which was discontinuous, causing the frequency to jump at certain critical temperatures. The X cut had a moderately bad coefficient, but it was more continuous, and by keeping the crystal in a temperature controlled oven, a high order of stability could be obtained. However, the X cut crystal was considerably less active than the Y cut, especially in the case of poorly ground plates. For frequencies between 500 kc. and about 6 Mc., the AT cut crystal now is the most widely used. It is active, can be made free from spurious responses, and has an excellent temperature characteristic. However, above about 6 Mc. it becomes quite thin, and a difficult production job. Between 6 Mc. and about 12 Mc., the BT cut plate is widely used. It also works well between 500 kc. and 6 Mc., but the AT cut is more desirable when a high order of stability is desired and no crystal oven is employed. For low frequency operation on the order of 100 kc., such as is required in a frequency standard, the GT cut crystal is recommended, though CT and DT cuts also are widely used for applications between 50 and 500 kc. The CT, DT, and GT cut plates are known as contour cuts, as these plates oscillate along the long dimension of the plate or bar, and are much smaller physically than would be the case for a regular AT or BT cut crystal for the same frequency. Crystal Holders Crystals normally are purchased ready mounted. The 244 Generation of R -F best type mount is determined by the type crystal and its application, and usually an optimum mounting is furnished with the crystal. However, certain features are desirable in all holders. One of these is exclusion of moisture and prevention of electrode oxidization. The best means of accomplishing this is a metal holder, hermetically sealed, with glass insulation and a metal -to -glass bond. However, such holders are more expensive, and a ceramic or phenolic holder with rubber gasket will serve where requirements are not too exacting. Temperature Control; Crystal Ovens Where the frequency tolerance requirements are not too stringent and the ambient temperature does not include extremes, an AT -cut plate, or a BT -cut plate with optimum (mean temperature) turning point, will often provide adequate stability without resorting to a temperature controlled oven. However, for broadcast stations and other applications where very close tolerances must be maintained, a thermostatically controlled oven, adjusted for a temperature slightly higher than the highest ambient likely to be encountered, must of necessity be employed. vibrating string can vibrate on its harmonics, a quartz crystal will Harmonic Cut Just as Crystals be made to THE Energy RADIO EXCITATION 6J5 ETC EXCITATION +5 loo -isov. BASIC PIERCE" OSCILLATOR HOT -CATHODE -PIERCE" OSCILLATOR Figure 6 THE PIERCE CRYSTAL OSCILLATOR CIRCUIT Shown at (A) is the basic Pierce crystal oscillator circuit. A capacitance of 10 to 75 µtd. normally will be required at C1 for optimum operation. If a plate supply voltage higher thon indicated is to be used, RFC' may be replaced by a 22,000 -ohm 2-watt resistor. Shown at (B) is an alternative arrangement with the r -f ground moved to the plate, and with the cathode floating. This alternative circuit has the advantage that the full r-f voltage developed across the crystal may be used os excitation to the next stage, since one side of the crystal is grounded. a exhibit mechanical résonance (and therefore electrical resonance) at harmonics of its funda- mental frequency. When employed in the usual holder, it is possible to excite the crystal only on its odd harmonics (overtones). By grinding the crystal especially for harmonic operation, it is possible to enhance its operation as a harmonic resonator. BT and AT cut crystals designed for optimum operation on the 3d, 5th and even the 7th harmonic are available. The 5th and 7th harmonic types, especially the latter, require special holder and oscillator circuit precautions for satisfactory operation, but the 3d harmonic type needs little more consideration than a regular fundamental type. A crystal ground for optimum operation on a particular harmonic may or may not be a good oscillator on a different harmonic or on the fundamental. One interesting characteristic of a harmonic cut crystal is that its harmonic frequency is not quite an exact multiple of its fundamental, though the disparity is very small. The harmonic frequency for which the crystal was designed is the working frequency. It is not the fundamental since the crystal itself actually oscillates on this working frequency when it is functioning in the proper manner. When a harmonic -cut crystal is employed, a selective tuned circuit must be employed somewhere in the oscillator in order to discrimi- nate against the fundamental frequency or undesired harmonics. Otherwise the crystal might not always oscillate on the intended frequency. For this reason the Pierce oscillator, later described in this chapter, is not suitable for use with harmonic -cut crystals, because the only tuned element in this oscillator circuit is the crystal itself. For a given crystal operating as an anti -resonant tank in a given oscillator at fixed load impedance and plate and screen voltages, the r -f current through the crystal will increase as the shunt capacitance C2 of figure 4 is increased, because this effectively increases the step -up ratio of C, to Cr. the crystal For a given shunt capacitance, current for a given crystal is directly proportional to the r -f voltage across C,. This voltage may be measured by means of a vacuum tube voltmeter having a low input capacitance, and such a measurement is a more pertinent one than a reading of r -f current by means of a thermogalvanometer inserted in series with one of the leads to the crystal holder. The function of a crystal is to provide accurate frequency control, and unless it is used in such a manner as to take advantage of its inherent high stability, there is no point in using a crystal oscillator. For this reason a Crystal Current; Heating and Fracture www.americanradiohistory.com C Crystal Oscillators HANDBOOK crystal oscillator should not be run at high plate input in an attempt to obtain considerable power directly out of the oscillator, as such operation will cause the crystal to heat, with resultant frequency drift and possible fracture. 13 -3 Crystal Oscillator Circuits Considerable confusion exists as to nomenclature of crystal oscillator circuits, due to a tendency to name a circuit after its discoverer. Nearly all the basic crystal oscillator circuits were either first used or else developed independently by G. W. Pierce, but he has not been so credited in all the literature. Use of the crystal oscillator in master oscillator circuits in radio transmitters dates back to about 1924 when the first application articles appeared. The Pierce The circuit of figure6A is the simplest crystal oscillator circuit. It Oscillator is one of those developed by Pierce, and is generally known among amateurs as the Pierce oscillator. The crystal simply replaces the tank circuit in a Colpitts or ultra-audion oscillator. The r -f excitation voltage available to the next stage is low, being somewhat less than that developed across the crystal. Capacitor C, will make more of the voltage across the crystal available for excitation, and sometimes will be found necessary to ensure oscillation. Its value is small, usually approximately equal to or slightly greater than the stray capacitance from the plate circuit to ground (including the grid of the stage being driven). If the r -f choke has adequate inductance, a crystal (even a harmonic cut crystal) will almost invariably oscillate on its fundamental. The Pierce oscillator therefore cannot be used with harmonic cut crystals. The circuit at (B) is the same as that of (A) except that the plate instead of the cathode is operated at ground r -f potential. All of the r -f voltage developed across the crystal is available for excitation to the next stage, but still is low for reasonable values of crystal current. For best operation a tube with low heater - cathode capacitance is required. Excitation for the next stage may also be taken from the cathode when using this circuit. The circuit shown in fig ure 7A is also one used by Pierce, but is more widely referred to as the " \filler "oscillator. To avoid Tuned -Plate Crystal Oscillator 245 confusion, we shall refer to it as the tuned plate crystal oscillator. It is essentially an Armstrong or tuned plate -tuned grid oscillator with the crystal replacing the usual L -C grid tank. The plate tank must be tuned to a frequency slightly higher than the anti -resonant (parallel resonant) frequency of the crystal. Whereas the Pierce circuits of figure 6 will oscillate at (or very close to) the anti -resonant frequency of the crystal, the circuits of figure 7 will oscillate at a frequency a little above the anti -resonant frequency of the crystal. The diagram shown in figure 7A is the basic circuit. The most popular version of the tuned plate oscillator employs a pentode or beam tetrode with cathode bias to prevent excessive plate dissipation when the circuit is not oscillating. The cathode resistor is optional. Its omission will reduce both crystal current and oscillator efficiency, resulting in somewhat more output for a given crystal current. The tube usually is an audio or video beam pentode or tetrode, the plate -grid capacitance of such tubes being sufficient to ensure stable oscillation but not so high as to offer excessive feedback with resulting high crystal current. The 6AG7 makes an excellent all- around tube for this type circuit. The usual type of crystal controlled h -f transmitter Oscillator Circuits operates, at least part of the time, on a frequency which is an integral multiple of the operating frequency of the controlling crystal. Hence, oscillator circuits which are capable of providing output on the crystal frequency if desired, but which also can deliver output energy on harmonics of the crystal frequency have come into wide use. Four such circuits which have found wide application are illustrated in figures 7C, 7D, 7E, and 7F. The circuit shown in figure 7C is recommended for use with harmonic -cut crystals when output is desired on a multiple of the oscillating frequency of the crystal. As an example, a 25-Mc. harmonic-cut crystal may be used in this circuit to obtain output on 50 Mc., or a 48 -Mc. harmonic-cut crystal may be used to obtain output on the 144-Mc. amateur band. The circuit is not recommended for use with the normal type of fundamental- frequency crystal since more output with fewer variable elements can be obtained with the circuits of 7D and 7F. The Pierce -harmonic circuit shown in figure 7D is satisfactory for many applications which require very low crystal current, but has the disadvantage that both sides of the crystal are above ground potential. The Tri -tet circuit of figure 7E is widely used and can Pentode Harmonic Crystal 246 Generation of 6J5,Erc F R -F 6V6. 6AQ5, RADIO THE Energy 6 AG7, 6AQ5. ETC 3F 5763 002 150V. +250 BASIC TUNED -PLATE OSCILLATOR 6AG7 F, 2F, 3F +250 V. V. SPECIAL C RCUIT FOR USE WITH HARMONIC CUT CRYSTAL. RECOMMENDED TUNED -PLATE OSCILLATOR 6AG7 F, 2F, 3F, 6AG7 4F F. 2F, 3F, 4F 10111JF 150 ULF 22 +250 K PIERCE HARMONIC CIRCUIT "TRITET" CIRCUIT Figure V. COLPITTS HARMONIC OSCILLATOR 7 COMMONLY USED CRYSTAL OSCILLATOR CIRCUITS Shown at (A) is the basic tuned -plate crystal oscillator with a triode oscillator tube. The plate tank must be tuned on the low -capacitance side of resonance to sustain oscillation. (8) shows the tuned -plat oscillator as it is normally used with an a -f power pentode to permit high output with relatively low crystal current. Schematics (C), (D), (E), and (F) illustrate crystal oscillator circuits which can deliver moderate output energy on harmonics of the oscillating frequency of the crystal. (C) shows a special circuit which will permit use of a harmonic -cut crystal to obtain output energy well into the v -h-f range. (D) is valuable when extremely low crystal current is a requirement, but delivers relatively low output. (E) is commonly used, but is subject to crystal damage if the cathode circuit is mistuned. (F) is recommended as the most generally satisfactory from the standpoints of: low crystal current regardless of mis- adjustment, good output on harmonic frequencies, one side of crystal is grounded, will oscillate with crystals from 1.5 to 70 Mc. without adjustment, output tank may be tuned to the crystal frequency for fundamental output without stopping oscillation or changing frequency. give excellent output with low crystal current. However, the circuit has the disadvantages of requiring a cathode coil, of requiring careful setting of the variable cathode capacitor to avoid damaging the crystal when changing frequency ranges, and of having both sides of the crystal above ground potential. The Colpitts harmonic oscillator of figure 7F is recommended as being the most generally satisfactory harmonic crystal oscillator circuit since it has the following advantages: (1) the circuit will oscillate with crystals over a very wide frequency range with no change other than plugging in or switching in the desired crystal; (2) crystal current is ex- tremely low; (3) one side of the crystal is grounded, which facilitates crystal- switching circuits; (4) the circuit will operate straight through without frequency pulling, or it may be operated with output on the second, third, or fourth harmonic of the crystal frequency. Crystal Oscillator Tuning The tunable circuits of all oscillators illustrated should be tuned for maximum output as indicated by maximum excitation to the following stage, except that the oscillator tank of tuned -plate oscillators (figure 7A and figure 7B) should be backed off slightly towards the low capacitance side from www.americanradiohistory.com HANDBOOK All -band Crystal Oscillator 6AG7 77. a 305 eaW M /N/JUC TOR (2 0 vN) +300 V. NOTES f" I. Li'/sUN (2 OF eew 0301s) (/" OF 80W .3003) 3. FOR 160 METER OPERATION ADD s 4/1/F. CONDENSER BETWEEN PINS 418 OF 6A67. PLATE COIL= 35 2/P+. 2. L2 = /.gLN (21-.0; Dew R awe) 4. X' 7 MC. CRYSTAL FOR HARMON /C OPERATION Figure 8 ALL -3AND 6AG7 CRYSTAL OSCILLATOR CAPABLE OF DRIVING BEAM PENTODE TUBE maximum output, as the oscillator then is in a more stable condition and sure to start immediately when power is applied. This is especially important when the oscillator is keyed, as for break-in c -w operation. It is desirable to keep stray shunt capacitances in the crystal circuit as low as possible, regardless of the oscillator circuit. If a selector switch is used, this means that both switch and crystal sockets must be placed close to the oscillator tube socket. This is especially true of harmonic -cut crystals operating on a comparatively high frequency. In fact, on the highest frequency crystals it is preferable to use a turret arrangement for switching, as the stray capacitances can be kept lower. Crystal Switching Crystol Oscillator When the crystal oscillator is keyed, it is necessary that crystal activity and oscillator -tube transconductance be moderately high, and that oscillator loading and crystal shunt capacitance be low. Below 2500 kc. and above 6 Mc. these considerations become especially important. Keying of the plate voltage (in the negative lead) of a crystal oscillator, with the screen voltage regulated at about 150 volts, has been found to give satisfactory results. Keying Versatile 6AG7 Crystal Oscillator A The 6AG7 tube may be used in a modified Tri -tet crystal oscillator, capable of delivering sufficient power on all bands 247 from 160 meters through 10 meters to fully drive a pentode tube, such as the 807, 2E26 or 6146. Such an oscillator is extremely useful for portable or mobile work, since it combines all essential exciter functions in one tube. The circuit of this oscillator is shown in figure 8. For 160, 80 and 40 meter operation the 6AG7 functions as a tuned-plate oscillator. Fundamental frequency crystals are used on these three bands. For 20, 15 and 10 meter operation the 6AG7 functions as a Tritet oscillator with a fixed -tuned cathode circuit. The impedance of this cathode circuit does not affect operation of the 6AG7 on the lower frequency bands so it is left in the circuit at all times. A 7 -Mc. crystal is used for fundamental output on 40 meters, and for harmonic output on 20, 15 and 10 meters. Crystal current is extremely low regardless of the output frequency of the oscillator. The plate circuit of the 6AG7 is capable of tuning a frequency range of 2:1, requiring only two output coils: one for 80-40 meter operation, and one for 20, 15 and 10 meter operation. In some cases it may be necessary to add 5 micromicrofarads of external feedback capacity between the plate and control grid of the 6AG7 tube to sustain oscillation with sluggish 160 meter crystals. Triode Overtone Oscillators The recent development of reliable overtone crystals capable of operation on the third, fifth, seventh (or higher) overtones has made possible v -h -f output from a low frequency crystal by the use of a double triode regenerative oscillator circuit. Some of the new twin triode tubes such as the 12AU7, 12AV7 and 6J6 are especially satisfactory when used in this type of circuit. Crystals that are ground for overtone service may be made to oscillate on odd overtone frequencies other than the one marked on the crystal holder. A 24 -Mc. overtone crystal, for example, is a specially ground 8 -Mc. crystal operating on its third overtone. In the proper circuit it may be made to oscillate on 40 Mc. (fifth overtone), 56 Mc. (seventh overtone), or 72 Mc. (ninth overtone). Even the ordinary 8 -Mc. crystals not designed for overtone operation may be made to oscillate readily on 24 Mc. (third overtone) in these circuits. A variety of overtone oscillator circuits is shown in figure 9. The oscillator of figure 9A is attributed to Frank Jones, W6AJF. The first section of the 6J6 dual triode comprises a regenerative oscillator, with output on either the third or fifth overtone of the crystal frequency. The regenerative loop of this oscillator consists of a condenser bridge made up of C, and C2, with the ratio C2 /C, determining the amount of regenerative feedback in the circuit. With www.americanradiohistory.com 248 Generation of R RADIO THE Energy -F 2F 6J6 4F 6,9,10 o915 F FOR 7 MC. CRYSTAL +300 +300 AO LI =28T e 24 ON NATIONAL V FORM L 2. $ T Il FORM V. © JONES HARMONIC OSCILLATOR 12AÚ7 XPSO ON NATIONAL XRSO COLPITTS HARMONIC OSCILLATOR 6J6 3F 6, 9F 3F /! /s0 50 6,9F RFC FOR +300 © v. L I= 9 7. L2. 4 T MC. CRYSTAL a 5003 9iW M /N /DUCTOR 13003 817 MIN/DOCTOR L +300V I = FOR 8MC. CRYSTAL M NIDUC TOR, TAP AT 37. FROM GRID ENO /07 a 30 /I 64W THESE COLS MADE FROM SINGLE SECT /ON OF NE TRN BROKEN TO O /V /OE IINDUCTOR INTO TWO COILS S REGENERATIVE HARMONIC OSCILLATOR OD REGENERATIVE HARMONIC OSCILLATOR - 12AT7 (or 6AB4) 12AT7 6,9F _ _ !00 F=IA4MC. 1_ L2 +200 LI=STe/E, L2=/ T. +300 E© V. f- O.SPACED NOOXUPWIRE, O. V. CATHODE FOLLOWER OVERTONE OSCILLATOR VARIOUS TYPES OF OVERTONE OF V.H.F. OVERTONE OSCILLATOR Figure 9 OSCILLATORS USING TUBES an 8 -plc. crystal, output from the first section of the 6J6 tube may be obtained on either 24 Mc. or 40 Mc., depending upon the resonant frequency of the plate circuit inductor, L,. The second half of the 636 acts as a frequency multiplier, its plate circuit, L2, tuned to the sixth or ninth harmonic frequency when L, is tuned to the third overtone, or to the tenth harmonic frequency when L, is tuned to the fifth overtone. Figure 9B illustrates a Colpitts overtone oscillator employing a 636 tube. This is an outgrowth of the Colpitts harmonic oscillator of figure 7F. The regenerative loop in this MINIATURE case consists of C DOUBLE -TRIODE C2 and RFC between the grid, cathode and ground of the first section of the 6J6. The plate circuit of the first section is tuned to the second overtone of the crystal, and the second section of the 636 doubles to the fourth harmonic of the crystal. This circuit is useful in obtaining 28 -Mc. output from a 7 -Mc. crystal and is highly popular in mobile work. The circuit of figure 9C shows a typical regenerative overtone oscillator employing a 12AÚ7 double triode tube. Feedback is controlled by the number of turns in L2, and the coupling between L2 and L,. Only enough feed- www.americanradiohistory.com HANDBOOK R back should be employed to maintain proper oscillation of the crystal. Excessive feedback will cause the first section of the 12ÁU7 to oscillate as a self- excited TNT oscillator, independent of the crystal. A variety of this circuit is shown in figure 9D, wherein a tapped coil, is used in place of the two separate coils. Operation of the circuit is the same in either case, regeneration now being controlled by the placement of the tap on L,. A cathode follower overtone oscillator is shown in figure 9E. The cathode coil, is chosen so as to resonate with the crystal and tube capacities just below the third overtone frequency of the crystal. For example, with an 8 -Mc. crystal, L3 is tuned to 24 Mc.. L, resonates with the circuit capacities to 23.5 Mc., and the harmonic tank circuit of the second section of the 12AT7 is tuned either to 48 Mc. or 72 11c. If a 24 -Mc. overtone crystal is used in this circuit, L, may be tuned to 72 Mc., L, resonates with the circuit capacities to 70 Mc., and the harmonic tank circuit, is tuned to 144 Mc. If there is any tendency towards self-oscillation in the circuit, it may be eliminated by a small amount of inductive coupling between L2 and L3. Placing these coils near each other, with the winding of L, correctly polarized with respect to L3 will prevent selfoscillation of the circuit. The use of a 144 -Mc. overtone crystal is illustrated in figure 9F. A 6AB4 or one -half of a 12AT7 tube may be used, with output directly in the 2 -meter amateur band. A slight amount of regeneration is provided by the one which is loosely coupled to the turn link, 144 -Mc. tuned tank circuit, L, in the plate circuit of the oscillator tube. If a 12AT7 tube and a 110 -Mc. crystal are employed, direct output in the 220 -Mc. amateur band may be obtained from the second half of the 12AT7. L L L L 13 -4 Radio Frequency Amplifiers The output of the oscillator stage in a transmitter (whether it be self-controlled or crystal controlled) must be kept down to a fairly low level to maintain stability and to maintain a factor of safety from fracture of the crystal when one is used. The low power output of the oscillator is brought up to the desired power level by means of radio -frequency amplifiers. The two classes of r -f amplifiers that find widest application in radio transmitters are the Class B and Class C types. The Class B Amplifier Class B amplifiers are used in a radio -telegraph transmitter when maximum power gain and mini- -F Amplifiers 249 output is desired in a particular stage. A Class B amplifier operates with cutoff bias and a comparatively small amount of excitation. Power gains of 20 to 200 or so are obtainable in a well- designed Class B amplifier. The plate efficiency of a Class B c -w amplifier will run around 65 per cent. mum harmonic Another type of Class B ampli fier is the Class B linear stage as employed in radiophone work. This type of amplifier is used to increase the level of a modulated carrier wave, and depends for its operation upon the linear relation between excitation voltage and output voltage. Or, to state the fact in another manner, the power output of a Class B linear stage varies linearly with the square of the excitation voltage. The Class B linear amplifier is operated with cutoff bias and a small value of excitation, the actual value of exciting power being such that the power output under carrier conditions is one -fourth of the peak power capabilities of the stage. Class B linears are very widely employed in broadcast and commercial installations, but are comparatively uncommon in amateur application, since tubes with high plate dissipation are required for moderate output. The carrier efficiency of such an amplifier will vary from approximately 30 per cent to 35 per cent. The Class B Linear Class C amplifiers are very wide ly used in all types of transmitters. Good power gain may be obtained (values of gain from 3 to 20 are common) and the plate circuit efficiency may be, under certain conditions, as high as 85 per cent. Class C amplifiers operate with considerably more than cutoff bias and ordinarily with a large amount of excitation as compared to a Class B amplifier. The bias for a normal Class C amplifier is such that plate current on the stage flows for approximately 120° of the 360° excitation cycle. Class C amplifiers are used in transmitters where a fairly large amount of excitation power is available and good plate circuit efficiency is desired. The Class C Amplifier The characteristic of a Class C amplifier which makes it linear with respect to changes in plate voltage is that which allows such an amplifier to be plate modulated for radiotelephony. Through the use of higher bias than is required for a c -w Class C amplifier and greater excitation, the linearity of such an amplifier may be extended from zero plate voltage to twice the normal value. The output power of a Class C amplifier, adjusted for plate modulation, varies with the square of the Plate Modulated Class C www.americanradiohistory.com Generation 250 of R -F THE Energy plate voltage. This is the same condition that would take place if a resistor equal to the voltage on the amplifier, divided by its plate current, were substituted for the amplifier. Therefore, the stage presents a resistive load RADIO Excessive grid current damages tubes by overheating the grid structure; beyond a certain point of grid drive, no increase in power output can be obtained for a given plate voltage. to the modulator. Grid Modulated If the grid current to a Class amplifier is reduced to a low value, and the plate loading is increased to the point where the plate dissipation approaches the rated value, such an amplifier may be grid modulated for radiotelephony. If the plate voltage is raised to quite a high value and the stage is adjusted carefully, efficiencies as high as 40 to 43 per cent with good modulation capability and comparatively low distortion may be obtained. Fixed bias is required. This type of operation is termed Class C grid-bias modulation. Class Grid Excitation Adequate grid excitation must be available for Class service. The excitation for a plate- modulated Class C stage must be sufficient to produce a normal value of d -c grid current with rated bias voltage. The bias voltage preferably should be obtained from a combination of grid leak and fixed C -bias supply. Cutoff bias can be calculated by dividing the amplification factor of the tube into the d-c plate voltage. This is the value normally used for Class B amplifiers (fixed bias, no grid resistor). Class C amplifiers use from 1 to 5 times this value, depending upon the available grid drive, or excitation, and the desired plate efficiency. Less grid excitation is needed for c -w operation, and the values of fixed bias (if greater than cutoff) may be reduced, or the value of the grid leak resistor can be lowered until normal rated d-c grid current flows. The values of grid excitation listed for each type of tube may be reduced by as much as 50 per cent if only moderate power output and plate efficiency are desired. When consulting the tube tables, it is well to remember that the power lost in the tuned circuits must be taken into consideration when calculating the available grid drive. At very high frequencies, the r -f circuit losses may even exceed the power required for actual grid excitation. Link coupling between stages, particularly to the final amplifier grid circuit, normally will provide more grid drive than can be obtained from other coupling systems. The number of turns in the coupling link, and the location of the turns on the coil, can be varied with respect to the tuned circuits to obtain the greatest grid drive for allowable values of buffer or doubler plate current. Slight readjustments sometimes can be made after plate voltage has been applied to the driver tube. B or Class Neutralization of R.F. Amplifiers 13 -5 C C C The plate -to -grid feedback capacitance of triodes makes it necessary that they be neutralized for operation as r -f amplifiers at frequencies above about 500 kc. Those screen grid tubes, pentodes, and beam tetrodes which have a plate -to -grid capacitance of 0.1 µµEd. or less may be operated as an amplifier without neutralization in a well -designed amplifier up to 30 Mc. Neutralizing Circuits The object of neutralization is to cancel or neutralize the capacitive feedback of energy from plate to grid. There are two general methods by which this energy feedback may be eliminated: the first, and the most common method, is through the use of a capacitance bridge, and the second method is through the use of a parallel reactance of equal and opposite polarity to the grid -to-plate capacitance, to nullify the effect of this capacitance. Examples of the first method are shown in figure 10. Figure l0A shows a capacity neutralized stage employing a balanced tank circuit. Phase reversal in the tank circuit is obtained by grounding the center of the tank coil to radio frequency energy by condenser C. Points A and B are 180 degrees out of phase with each other, and the correct amount of out of phase energy is coupled through the neutralizing condenser NC to the grid circuit of the tube. The equivalent bridge circuit of this is shown in figure 11A. It is seen that the bridge is not in balance, since the plate -filament capacity of the tube forms one leg of the bridge, and there is no corresponding capacity from the neutralizing condenser (point B) to ground to obtain a complete balance. In addition, it is mechanically difficult to obtain a perfect electrical balance in the tank coil, and the potential between point A and ground and point B and ground in most cases is unequal. This circuit, therefore, holds neutralization over a very small operating range and unless tubes of low interelectrode capacity are used the inherent unbalance of the circuit will permit only approximate neutralization. Split-Stator Plate Neutralization www.americanradiohistory.com Figure 10B shows the neu- tralization circuit which is most widely used in single ended r -f stages. The use of HANDBOOK Neutralization Figure COMMON 10 NEUTRALIZING CIRCUITS FOR SINGLE -ENDED AMPLIFIERS split -stator plate capacitor makes the electrical balance of the circuit substantially independent of the mutual coupling within the coil and also makes the balance independent of the place where the coil is tapped. With conventional tubes this circuit will allow one neutralization adjustment to be made on, say, 28 Mc., and this adjustment usually will hold sufficiently close for operation on all lower frequency bands. Condenser C, is used to balance out the plate -filament capacity of the tube to allow a perfect neutralizing balance at all frequencies. The equivalent bridge circuit is shown in figure 11B. If the plate- filament capacity of the tube is extremely low (100TH triode, for example), condenser C, may be omitted, or may merely consist of the residual capacity of NC to ground. a split grid tank circuit also be used for neutralization of a triode tube as shown in figure IOC. Out of phase voltage is developed across a balanced grid circuit, and coupled through NC to the single -ended plate circuit of the tube. The equivalent bridge circuit is shown in figure 11C. This circuit is in balance until the stage is in operation when the loading effect of the tube upon one -half of the grid circuit throws the bridge circuit out of balance. The amount of unbalance depends upon the grid -plate capacity of the tube, and the amount of mutual inductance between the two halves Grid Neutralization 251 A may of the grid coil. If an r -f voltmeter is placed between point A and ground, and a second voltmeter placed between point B and ground the loading effect of the tube will be noticeable. When the tube is supplied excitation with no plate voltage, NC may be adjusted until the circuit is in balance. When plate voltage is applied to the stage, the voltage from point A to ground will decrease, and the voltage from point B to ground will increase, both in direct proportion to the amount of circuit unbalance. The use of this circuit is not recommended above 7 Mc., and it should be used below that frequency only with low in- ternal capacity tubes. Two tubes of the same type can be connected for push -pull operation so as to obtain twice as much output as that of a single tube. A push -pull amplifier, such as that shown in figure 12 also has an advantage in that the circuit can more easily be balanced than a single tube r -f amplifier. The various inter -electrode capacitances and the neutralizing capacitors are connected in such a manner that the reactances on one side of the tuned circuits are exactly equal to those on the opposite side. For this reason, push -pull r -f amplifiers can be more easily neutralized in very- high -frequency transmitters; also, they usually remain in perfect neutralization when tuning the amplifier to different bands. The circuit shown in figure 12 is perhaps Push -Pull Neutralization Generation of 252 -F R THE Energy RADIO F.AC OA BRIDGE EQUIVALENT OF FIGURE IO -A C Figure 12 STANDARD CROSS- NEUTRALIZED PUSH -PULL TRIODE AMPLIFIER OB BRIDGE EQUIVALENT OF FIGURE 10 -B C actance, coupling energy back from the plate to the grid circuit. If this capacitance is paralleled with an inductance having the same value of reactance of opposite sign, the reactance of one will cancel the reactance of the other and a high -impedance tuned circuit will result. This neutralization circuit can be used on ultra -high frequencies where other neutralization circuits are unsatisfactory. This is true because the lead length in the neutralization circuit is practically negligible. The circuit can also be used with push -pull r -f amplifiers. In this case, each tube will have its own neutralizing inductor connected from grid to plate. The main advantage of this arrangement is that it allows the use of single -ended tank circuits with a single -ended amplifier. The chief disadvantage of the shunt neutralized arrangement is that the stage must be reneutralized each time the stage is retuned to a new frequency sufficiently removed that the grid and plate tank circuits must be retuned to resonance. However, by the use of plug -in coils it is possible to change to a different band of operation by changing the neutralizing coil at the same time that the grid and plate coils are changed. The 0.0001-pfd. capacitor in series with the neutralizing coil is merely a blocking capacitor to isolate the plate voltage from the grid circuit. The coil L will have to have a very large number of turns for the band of operation in order to be resonant with the comparatively small grid -to -plate capacitance. But since, in all ordinary cases with tubes operating on frequencies for which they were designed, the L/C ratio of the tuned circuit will be very high, the coil can use comparatively small wire, although it must be wound on air or very low loss dielectric and must be insulated for the sum of the plate r-f voltage and the grid r-f voltage. from grid to plate (RES.DUAL CAPACITY)` © CG-r (s1IALL) RFC BRIDGE EQUIVALENT OF FIGURE 10-G Figure 11 EQUIVALENT NEUTRALIZING CIRCUITS most commonly used arrangement for a push -pull r -f amplifier stage. The rotor of the grid capacitor is grounded, and the rotor of the plate tank capacitor is by- passed to ground. the Shunt or Coil Neutralization The feedback of energy from grid to plate in an unneutral- ized r -f amplifier is a result of the grid -to -plate capacitance of the amplifier tube. A neutralization circuit is merely an electrical arrangement for nullifying the effect of this capacitance. All the previous neutralization circuits have made use of a bridge circuit for balancing out the grid -to -plate energy feedback by feeding hack an equal amount of energy of opposite phase. Another method of eliminating the feedback effect of this capacitance, and hence of neutralizing the amplifier stage, is shown in figure 13. The grid -to -plate capacitance in the triode amplifier tube acts as a capacitive re- www.americanradiohistory.com Neutralizing HANDBOOK Procedure 253 grid excitation is applied, even though no primary a-c voltage is being fed to the plate transformer. further check on the neutralization of any amplifier can be made by noting whether maximum grid current on the stage comes at the same point of tuning on the plate tuning capacitor as minimum plate current. This check is made with plate voltage on the amplifier and with normal antenna coupling. As the plate tuning capacitor is detuned slightly from resonance on either side the grid current on the A r -f Figure 13 COIL NEUTRALIZED AMPLIFIER This neutralization circuit is very effective with triode tubes on any frequency, but is particularly effective in the v -h-f range. The coil L is adjusted so that it resonates at the operating frequency with the grid -to -plate capacitance of the tube. Capacitor C may be a very small unit of the low- capacitance neutralizing type and is used to trim the circuit to resonance at the operating frequency. If some means of varying the inductance of the coil a small amount is available, the trimmer capacitor is not needed. stage should decrease the same amount and without any sudden jumps on either side of resonance. This will be found to be a very precise indication of accurate neutralization in either a triode or beam -tetrode r -f amplifier stage, so long as the stage is feeding a load which presents a resistive impedance at the operating frequency. Push-pull circuits usually can be more completely neutralized than single -ended circuits at very high frequencies. In the intermediate range of from 3 to 15 Mc., single -ended circuits will give satisfactory results. Neutralization of Screen -Grid 13 -6 R -F Amplifiers Neutralizing Procedure An r-f amplifier is neutralized to prevent self-oscillation or regeneration. A neon bulb, a flashlight lamp and loop of wire, or an r -f galvanometer can be used as a null indicator for neutralizing low -power stages. The plate voltage lead is disconnected from the r-f amplifier stage while it is being neutralized. Normal grid drive then is applied to the r -f stage, the neutralizing indicator is coupled to the plate coil, and the plate tuning capacitor is tuned to resonance. The neutralizing capacitor (or capacitors) then can be adjusted until minimum r.f. is indicated for resonant settings of both grid and plate tuning capacitors. Both neutralizing capacitors are adjusted simultaneously and to approximately the same value of capacitance when a physically symmetrical push -pull stage is being neutralized. A final check for neutralization should be made with a d-c milliammeter connected in the grid leak or grid -bias circuit. There will be no movement of the meter reading as the plate circuit is tuned through resonance (without plate voltage being applied) when the stage is completely neutralized. Plate voltage should be completely removed by actually opening the d-c plate circuit. If there is a d-c return through the plate supply, a small amount of plate current will flow when Radio-frequency amplifiers using screen -grid tubes can be operated without any ad- ditional provision for neutralization at frequencies up to about 15 Mc., provided adequate shielding has been provided between the input and output circuits. Special v -h -f screen -grid and beam tetrode tubes such as the 2E26, 807W, and 5516 in the low -power category and HK -257B, 4E27/8001, 4 -125A, and 4 -250A in the medium -power category can frequently be operated at frequencies as high as 100 Mc. without any additional provision for neutralization. Tubes such as the 807, 2E22, HY -69, and 813 can be operated with good circuit design at frequencies up to 30 Mc. without any additional provision for neutralization. The 815 tube has been found to require neutralization in many cases above 20 Mc., although the 829B tube will operate quite stably at 100 Mc. without neutralization. None of these tubes, however, has perfect shielding between the grid and the plate, a condition brought about by the inherent inductance of the screen leads within the tube itself. In addition, unless "watertight" shielding is used between the grid and plate circuits of the tube a certain amount of external leakage between the two circuits is present. These difficulties may not be serious enough to require neutralization of the stage to prevent oscillation, but in many instances they show up in terms of key -clicks when the stage in question is keyed, or as parasitics when the stage is modulated. Unless the designer of the equipment can carefully check the tetrode www.americanradiohistory.com 254 Generation of R -F THE Energy Figure RADIO 14 NEUTRALIZING CIRCUITS FOR BEAM TETRODES conventional cross neutralized circuit for use with push-pull beam tetrodes is shown at (A). The neutralizing capacitors (NC) usually consist of small plates or rods mounted alongside the plate elements of the tubes. (B) and (C) show "grid neutralized" circuits for use with a single -ended tetrode stage having either link coupling or capacitive coupling into the grid tank. (D) shows a method of tuning the screen -lead inductance to accomplish neutralization in a single -frequency -h-f tetrode amplifier, while (E) shows a method of neutralization by increasing the vgrid to -plate capacitance on a tetrode when the operating frequency is higher than that frequency where the tetrode is "self- neutralized" as a result of series resonance in the screen lead. Methods (D) and (E) normally are not practicable at frequencies below about 50 Mc. with the usual types of beam tetrode tubes. A stage for miscellaneous feedback between the grid and plate circuits, and make the necessary circuit revisions to reduce this feedback to an absolute minimum, it is wise to neutralize the tetrode just as if it were a triode tube. In most push -pull tetrode amplifiers the simplest method of accomplishing neutralization is to use the cross -neutralized capacitance bridge arrangement as normally employed with triode tubes. The neutralizing capacitances, however, must be very much smaller than used with triode tubes, values of the order of 0.2 wifd. normally being required with beam tetrode tubes. This order of capacitance is far less than can be obtained with a conventional neutralizing capacitor at minimum setting, so the neutralizing arrangement is most commonly made especially for the case at hand. Most common procedure is to bring a conductor (connected to the opposite grid) in the vicinity of the plate itself or of the plate tuning capacitor of one of the tubes. Either one or two such capacitors may be used, two being normally used on a higher frequency amplifier in order to maintain balance within the stage. An example of this is shown in figure 14A. www.americanradiohistory.com H Tetrode Neutralization A N D B O O K Neutralizing single -ended tetrode r -f amplifier stage may be neutral ized in the same manner as illustrated for a push -pull stage in figure 14A, provided a split- stator tank capacitor is in use in the plate circuit. However, in the majority of single -ended tetrode r -f amplifier stages a single- section capacitor is used in the plate tank. Hence, other neutralization procedures must be employed when neutralization is found necessary. The circuit shown in figure 14B is not a true neutralizing circuit, in that the plate -togrid capacitance is not balanced out. However, the circuit can afford the equivalent effect by isolating the high resonant impedance of the grid tank circuit from the energy fed back from plate to grid. When NC and C are adjusted to bear the following ratio to the grid -to-plate capacitance and the total capacitance from A Single -Ended Tetrode Stages grid -to- ground in the output tube: NC CsP C Cas both ends of the grid tank circuit will be at the same voltage with respect to ground as a result of r -f energy fed back to the grid circuit. This means that the impedance from grid to ground will be effectively equal to the reactance of the grid -to- cathode capacitance in parallel with the stray grid -to- ground capacitance, since the high resonant impedance of the tuned circuit in the grid has been effectively isolated from the feedback path. It is important to note that the effective grid -to- ground capacitance of the tube being neutralized includes the rated grid -to- cathode or input capacitance of the tube, the capacitance of the socket, wiring capacitances and other strays, but it does not include the capacitances associated with the grid tuning capacitor. Also, if the tube is being excited by capacitive coupling from a preceding stage (as in figure 14C), the effective grid-to- ground capacitance includes the output capacitance of the preceding stage and its associated socket and wiring capacitances. The provisions discussed in the previous paragraphs are for neutralization of the small, though still important at the higher frequencies, grid -to -plate capacitance of beam -tetrode tubes. However, in the vicinity of the upper- frequency limit of each tube type the inductance of the screen lead of the tube becomes of considerable importance. With a tube operating at a frequency where the inductance of the screen lead is appreciable, the screen will allow a considerable amount of energy leak- through from plate to grid even Cancellation of Screen -Lead Inductance 255 though the socket terminal on the tube is carefully by- passed to ground. This condition takes place even though the socket pin is bypassed since the reactance of the screen lead will allow a moderate amount of r-f potential to appear on the screen itself inside the electrode assembly in the tube. This effect has been reduced to a very low amount in such tubes as the Hytron 5516, and the Eimac 4X150A and 4X500A but it is still quite appreciable in most beam -tetrode tubes. The effect of screen -lead inductance on the stability of a stage can be eliminated at any particular frequency by one of two methods. These methods are: (1) Tuning out the screen lead inductance by series resonating the screen lead inductance with a capacitor to ground. This method is illustrated in figure 14D and is commonly employed in commercially -built equipment for operation on a narrow frequency band in the range above about 75 Mc. The other method (2) is illustrated in figure 14E and consists in feeding back additional energy from plate to grid by means of a small capacitor connected between these two elements. Note that this capacitor is connected in such a manner as to increase the effective grid -toplate capacitance of the tube. This method has been found to be effective with 807 tubes in the range above 50 Mc. and with tubes such as the 4 -125A and 4 -250A in the vicinity of their upper frequency limits. Note that both these methods of stabilizing a beam-tetrode v -h -f amplifier stage by cancellation of screen -lead inductance are suitable only for operation over a relatively narrow band of frequencies in the v -h-f range. At lower frequencies both these expedients for reducing the effects of screen -lead inductance will tend to increase the tendency toward oscillation of the amplifier stage. stage cannot be completely neutralized, the difficulty usually can be traced to one or more of the following causes: (1) Filament leads not by- passed to the common ground of that particular stage. (2) Ground lead from the rotor connection of the split -stator tuning capacitor to filament open or too long. (3) Neutralizing capacitors in a field of excessive r.f. from one of the tuning coils. (4) Electromagnetic coupling between grid and plate coils, or between plate and preceding buffer or oscillator circuits. (5) Insufficient shielding or spacing between stages, or between grid and plate circuits in compact transmitters. (6) Shielding placed too close to plate circuit coils, causing induced currents in the shields. (7) Parasitic oscillations when plate voltage is applied. The cure for the latter is mainly a matter of cut and try -rearrange the parts, Neutralizing Problems When a - 256 Generation of R -F THE Energy GRID LEAK RADIO OUT INTERWOUND COILS (UNITY COUPLING) Figure CONVENTIONAL 16 TRIODE FREQUENCY MULTIPLIER Figure 15 GROUNDED -GRID AMPLIFIER This type of triode amplifier requires no neutralization, but can be used only with tubes having o relatively low plate -to- cathode capacitance change the length of grid or plate or neutralizing leads, insert a parasitic choke in the grid lead or leads, or eliminate the grid r -f chokes which may be the cause of a low- frequency Small triodes such as the 604 operate satisfactorily as frequency multipliers, and can deliver output well into the v -h -t ronge. Resistor R normally will have a value in the vicinity of 100,000 ohms. given output, because a moderate amount of power is delivered to the amplifier load by the driver stage of a grounded -grid amplifier. 13 -8 Frequency Multipliers parasitic(in conjunction with plate r -f chokes). 13- 7 Grounded Grid Amplifiers Certain triodes have grid configuration results in very low plate to filament capacitance when the control grid is grounded, the grid acting as an effective shield much in the manner of the screen in a screen -grid tube. By connecting such a triode in the circuit of figure 15, taking the usual precautions against stray capacitive and inductive coupling between input and output leads and components, a stable power amplifier is realized which requires no neutralization. At ultra -high frequencies, where it is difficult to obtain satisfactory neutralization with conventional triode circuits (particularly when a wide band of frequencies is to be covered), the grounded -grid arrangement is about the only practicable means of employing a triode ama and lead arrangement which plifier. Because of the large amount of degeneration inherent in the circuit, considerably more excitation is required than if the same tube were employed in a conventional grounded- cathode circuit. The additional power required to drive a triode in a grounded -grid amplifier is not lost, however, as it shows up in the output circuit and adds to the power delivered to the load. But nevertheless it means that a larger driver stage is required for an amplifier of Quartz crystals and variable- frequency oscillators are not ordinarily used for direct control of the output of high- frequency transmitters. Frequency multipliers are usually employed to multiply the frequency to the desired value. These multipliers operate on exact multiples of the excitation frequency; a 3.6-Mc. crystal oscillator can be made to control the output of a transmitter on 7.2 or 14.4 Mc., or on 28.8 Mc., by means of one or more frequency multipliers. Chen used at twice frequency, they are often termed frequency doublers. A simple doubler circuit is shown in figure 16. It consists of a vacuum tube with its plate circuit tuned to twice the frequency of the grid driving circuit. This doubler can be excited from a crystal oscillator or another multiplier or amplifier stage. Doubling is best accomplished by operating the tube with high grid bias. The grid circuit is driven approximately to the normal value of d -c grid current through the r-f choke and grid leak resistor, shown in figure 16. The resistance value generally is from two to five times as high as that used with the same tube for straight amplification. Consequently, the grid bias is several times as high for the same value of grid current. Neutralization is seldom necessary in a doubler circuit, since the plate is tuned to twice the frequency of the grid circuit. The impedance of the grid driving circuit is very low at the doubling frequency, and thus there is little tendency for self -excited oscillation. www.americanradiohistory.com HANDBOOK Frequency Multipliers 257 TANN CIRCUIT OUTPUT VOLTAGE s I (CUTOrr) n A A \f/ \I A N C,, K GI I,J N \ L1 M O A á A \\ U1 I 1 W T 1 N(cuTOn)------¡--P--t-- i - - -1 \ \ i EXCITATION VOLTAGE \O Figure 18 ILLUSTRATING THE ACTION OF A FREQUENCY DOUBLER degrees or less. Under these conditions the efficiency will be on the same order as the reciprocal of the harmonic on which the stage operates. In other words the efficiency of a doubler will be approximately % or 50 per cent, the efficiency of a tripler will be approximately / or 33 per cent and that of a quadrupler will be about 25 per cent. With good stage design the efficiency can be somewhat greater than these values, but as the angle of flow is made greater than these limiting values, the efficiency falls off rapidly. The reason is apparent from a study of figure 18. The pulses ABC, EFG, JKL illustrate 180 degree excitation pulses under Class B operation, the solid straight line indicating cutoff bias. If the bias is increased by N times, to the value indicated by the dotted straight line, and the excitation increased until the peak r -f voltage with respect to ground is the same as before, then the excitation frequency can be cut in half and the effective excitation pulses will have almost the same shape as before. The only difference is that every other pulse is missing; MNO simply shows where the missing pulse would go. However, if the Q of the plate tank circuit is high, it will have sufficient flywheel effect to carry over through the missing pulse, and the only effect will be that the plate input and r -f output at optimum loading drop to approximately half. As the input frequency is half the output frequency, an efficient frequency doubler is the result. By the same token, a tripler or quadrupler can be analyzed, the tripler skipping two excitation pulses and the quadrupler three. In each case the excitation pulse ideally should be short enough that it does not exceed 180 degrees at the output frequency; otherwise the excitation actually is bucking the output over a portion of the cycle. In actual practice, it is found uneconomical to provide sufficient excitation to run a tripler or quadrupler in this fashion. Usually the ex45 Figure 17 FREQUENCY MULTIPLIER CIRCUITS The output of a triode v-h -f frequency multiplier often maybe increased by neutralization of the grid-to -plate capacitance as shown at (A) above. Such o stage also may be operated as a straight amplifier when the occasion demands. A pentode frequency multiplier is shown at (B). Conventional power tetrodes operate satisfactorily as multipliers so long as the output frequency is below about 100 Mc. Above this frequency special v -h -f tetrodes must be used to obtain satisfactory output. Frequency doublers require bias of several times cutoff; high -ft tubes therefore are desirable for this type of service. Tubes which have amplification factors from 20 to 200 are suitable for doubler circuits. Tetrodes and pentodes make excellent doublers. Low -ft triodes, having amplification constants of from 3 to 10, are not applicable for doubler service. In extreme cases the grid voltage must be as high as the plate voltage for efficient doubling action. Angle of Flow in Frequency The angle of plate current flow in a frequency multiplier is a Multipliers very important factor in determining the efficiency. As the angle of flow is decreased for a given value of grid current, the efficiency increases. To reduce the angle of flow, higher grid bias is required so that the grid excitation voltage will exceed the cutoff value for a shorter portion of the exciting -voltage cycle. For a high order of efficiency, frequency doublers should have an angle of flow of 90 degrees or less, tripiers 60 degrees or less, and quadruplers www.americanradiohistory.com Generation of 258 Figure R -F THE Energy RADIO 19 Figure PUSH -PUSH FREQUENCY DOUBLER The output of o doubler stage may be materially increased through the use of a push -push circuit such as illustrated above. citation pulses will be at least 90 degrees at the exciting frequency, with correspondingly low efficiency, but it is more practicable to accept the low efficiency and build up the output in succeeding amplifier stages. The efficiency can become quite low before the power gain becomes less than unity. Two tubes can be connected in parallel to give twice the output of a single -tube doubler. If the grids are driven out of phase instead of in phase, the tubes then no longer work simultaneously, but rather one at a time. The effect is to fill in the missing pulses (figure 18). Not only is the output doubled, but several advantages accrue which cannot be obtained by straight parallel operation. Chief among these is the effective neutralization of the fundamental and all odd harmonics, an advantage when spurious emissions must be minimized. Another advantage is that when the available excitation is low and excitation pulses exceed 90 degrees, the output and efficiency will be greater than for the same tubes connected in parallel. The same arrangement may be used as a quadrupler, with considerably better efficiency than for straight parallel operation, because seldom is it practicable to supply sufficient excitation to permit 45 degree excitation pulses. As pointed out above, the push -push arrangement exhibits better efficiency than a single ended multiplier when excitation is inadequate for ideal multiplier operation. A typical push -push doubler is illustrated in figure 19. When high transconductance tubes are employed, it is necessary to employ a split- stator grid tank capacitor to prevent self oscillation; with well screened tetrodes or pentodes having medium values of transconductance, a split -coil arrangement with a single- section capacitor may be employed (the 20 PUSH -PULL FREQUENCY TRIPLER The push -pull tripler is advantageous in the v -h-f ronge since circuit balance is maintained both in the input and output circuits. If the circuit is neutralized it may be used either as a straight amplifier or as a tripler. Either triodes or tetrodes may be used; dual unit tetrodes such as the 815, 832A, and 8298 are particularly effective in the v-h -f range. center tap of the grid coil being by- passed to ground). Push -Push Multipliers Push -Pull Frequency It is frequently desirable in the case of u -h -f and v -h -f transmitters that frequency multiplication stages be balanced with respect to ground. Further it is just as easy in most cases to multiply the crystal or v -f -o frequency by powers of three rather than multiplying by powers of two as is frequently done on lower frequency transmitters. Hence the use of push -pull tripiers has become quite prevalent in both commercial and amateur v -h -f and u -h -f transmitter designs. Such stages are balanced with respect to ground and appear in construction and on paper essentially the same as a push -pull r -f amplifier stage with the exception that the output tank circuit is tuned to three times the frequency of the grid tank circuit. A circuit for a push -pull tripler stage is shown in figure 20. A push -pull tripler stage has the further advantage in amateur work that it can also be used as a conventional push -pull r -f amplifier merely by changing the grid and plate coils so that they tune to the same frequency. This is of some advantage in the case of operating the 50 -Mc. band with 50 -bic. excitation, and then changing the plate coil to tune to 144 Mc. for operation of the stage as a tripler from excitation on 48 Mc. This circuit arrangement is excellent for operation with push -pull beam tetrodes such as the 6360 and 829B, although a pair of tubes such as the 2E26, or 5763 could just as well be used if proper attention were given to the matter of screen -lead inductance. Tripiers www.americanradiohistory.com Tank HANDBOOK Tank Circuit Capacitances 13 -9 It is necessary that the proper value of Q plate tank circuit of any r -f amplifier. The following section has been devoted to a treatment of the subject, and charts are given to assist the reader in the determination of the proper L/C ratio to be used in a radio -frequency amplifier stage. A Class C amplifier draws plate current in the form of very distorted pulses of short duration. Such an amplifier is always operated into a tuned inductance- capacitance or tank circuit which tends to smooth out these pulses, by its storage or tank action, into a sine wave DYNAMIC CHARACTERISTIC Tank Circuit Q As stated before, the tank cir- cuit of a Class C amplifier receives energy in the form of short pulses of plate current which flow in the amplifier tube. But the tank circuit must be able to store enough energy so that it can deliver a current essentially sine wave in form to the load. The ability of a tank to store energy in this manner may be designated as the effective Q of the tank circuit. The effective circuit Q may be stated in any of several ways, but essentially the Q of a tank circuit is the ratio of the energy stored to 2e times the energy lost per cycle. Further, the energy lost per cycle must, by definition, be equal to the energy delivered to the tank circuit by the Class C amplifier tube or tubes. The Q of a tank circuit at resonance is equal to its parallel resonant impedance (the resonant impedance is resistive at resonance) divided by the reactance of either the capacitor or the inductor which go to make up the tank. The inductive reactance is equal to the capacitive reactance, by definition, at resonance. Hence we may state: 259 A_ (0\ be used in the of radio -frequency output. Any wave -form distortion of the carrier frequency results in harmonic interference in higher- frequency channels. A Class A r-f amplifier would produce a sine wave of radio -frequency output if its exciting waveform were also a sine wave. However, a Class A amplifier stage converts its d -c input to r -f output by acting as a variable resistance, and therefore heats considerably. A Class C amplifier when driven hard with short pulses at the peak of the exciting waveform acts more as an electronic switch, and therefore can convert its d -c input to r -f output with relatively good efficiency. Values of plate circuit efficiency from 65 to 85 per cent are common in Class C amplifiers operating under optimum conditions of excitation, grid bias, and loading. Circuits GRID SWING Figure 21 AMPLIFIER OPERATION Plate current pulses are shown at (A), (e), and (C). The dip in the top of the plate current waveform will occur when the excitation voltage is such that the minimum plate voltage dips below the maximum grid voltage. A detailed discussion of the operation of Class C amplifiers is given in Chapter Seven. CLASS C Q = -=RL RL Xc XL where RL is the resonant impedance of the tank and Xc is the reactance of the tank capacitor and XL is the reactance of the tank coil. This value of resonant impedance, RL, is the load which is presented to the Class C amplifier tube in a single -ended circuit such as shown in figure 21. The value of load impedance, RL, which the Class C amplifier tube sees may be obtained, looking in the other direction from the tank coil, from a knowledge of the operating conditions on the Class C tube. This load impedance may be obtained from the following expression, which is true in the general case of any Class C amplifier: Epm= RL 2 Np lb Ebb where the values in the equation have the characteristics listed in the beginningof Chapter 6. The expression above is academic, since the peak value of the fundamental component of plate voltage swing, Epm, is not ordinarily known unless a high -voltage peak a-c voltmeter is available for checking. Also, the decimal value of plate circuit efficiency is not ordinarily known with any degree of accuracy. However, in a normally operated Class C amplifier Generation of 260 R > s w > ! 54 V- 8 3. u x 100 ,S 10 TANK CIRCUIT 30 25 20 Q THE Energy -F which means simply that the resistance presented by the tank circuit to the Class C tube is approximately equal to one -half the d -c load resistance which the Class C stage presents to the power supply (and also to the modulator in case high -level modulation of the stage is to be used). Combining the above simplified expression for the r -f impedance presented by the tank to the tube, with the expression for tank Q given in a previous paragraph we have the following expression which relates the reactance of the tank capacitor or coil to the d-c input to the Class C stage: XC Figure 22 RELATIVE HARMONIC OUTPUT PLOTTED AGAINST TANK CIRCUIT .v Rd. c. \\\\1 \ \ II 2 20000 15 10 i Ó > w I- XL , Rd.c. The above expression is the basis of the usual charts giving tank capacitance for the various bands in terms of the d -c plate voltage and current to the Class C stage, including the charts of figure 23, figure 24 and figure 25. Harmonic Rodio- The problem of harmonic radiation from transmitters has long been present, but it has become critical only relatively recently along with the extensive occupation of the v -h -f range. Television signals are particularly susceptible to interference from other signals falling within the pass band of the receiver, so that the TVI problem has received the major emphasis of all the services in the v -h -f range which are susceptible to interference from harmonics of signals in the h -f or lower v -h -f range. tion vs. Q \11111III1N1 111111!\IIIIIII \\ Q=12 II w = 2Q Q the plate voltage swing will be approximately equal to 0.85 to 0.9 times the d -c plate voltage on the stage, and the plate circuit efficiency will be from 70 to 80 per cent (Np of 0.7 to 0.8), the higher values of efficiency normally being associated with the higher values of plate voltage swing. With these two assumptions as to the normal Class C amplifier, the expression for the plate load impedance can be greatly simplified to the following approximate but useful expression: RL RADIO III IIII mum 111 . 11 1 NEUTRALIZING COIL \11'I \\111Ii 1111, I\IIII (I RFC -e 1110M1121111101111111111 3 10 20 IIII!ÍiNHO!i 30 100 500 200 TOTAL CAPACITANCE ACROSS LC C 1000 2000 RCUIT (CO O Figure 23 PLATE -TANK CIRCUIT ARRANGEMENTS Shown above in the case of each of the tank circuit types is the recommended tank circuit capacitance. (A) is a conventional tetrode amplifier, (B) is a coil -neutralized triode amplifier, (C) is a grounded-grid triode amplifier, (D) is a grid -neutralized triode amplifier. www.americanradiohistory.com re HANDBOOK 10 Tank Circuits 261 \.,'..O..U...n \/ \11 III IIII ÌIÌI!I\1iIÌCIÌI\\IIIIIÌ MONIIIMMIIIMUM111 \\111i111\11111111111111 1MINE111011; I111111MII1,vIM11111MMnII 1.3 o > á II ¢ 1 11 1 1 41 1 11 1 1 ., IlliliE1111P11 1111111 et 11111111111111111111 10 2 3 5 7 lo 30 50 100 lao 500 1000 CORRECT VALUES OF TANK CIRCUIT CAPACITANCE (C FOR OPERATING Q OF 12 WITH SINGLE-ENDED SPLIT TANK COILS Figure 24 PLATE -TANK CIRCUIT ARRANGEMENTS Shown above for each of the tank circuit types is the recommended tank circuit capacitance of the operating frequency for an operating Q of 12. (A) is a split -stator tank, each section of which is twice the capacity value read on the graph. (8) is circuit using tapped coil for phase reversal. Inspection of figure 22 will show quickly that the tank circuit of a Class C amplifier should have an operating Q of 12 or greater to afford satisfactory rejection of second harmonic energy. The curve begins to straighten out above a Q of about 15, so that a considerable increase in Q must be made before an appreciable reduction in second -harmonic energy is obtained. Above a circuit Q of about 10 any increase will not afford appreciable reduction in the third -harmonic energy, so that additional harmonic filtering circuits external to the amplifier proper must be used if increased attenuation of higher order harmonics is desired. The curves also show that push -pull amplifiers may be operated at Q values of 6 or so, since the second harmonic is cancelled to a large extent if there is no unbalanced coupling between the output tank circuit and the antenna system. Figures 23, 24 and 25 illustrate the correct value of tank capacity for various circuit configurations. A Q value of 12 has been chosen as optimum for single ended circuits, and a value of 6 has been chosen for push -pull circuits. Figure 23 is used when a single ended stage is employed, and the capacitance values given are for the total capacitance across the tank coil. This value includes the tube interelectrode capacitance (plate to ground), coil distributed capacitance, wiring capacities, and the value of any lowCapacity Charts for Correct Tank Q inductance plate -to- ground by -pass capacitor as used for reducing harmonic generation, in addition to the actual "in -use" capacitance of the plate tuning capacitor. Total circuit stray capacitance may vary from perhaps 5 micromicrofarads for a v -h -f stage to 30 micro microfarads for a medium power tetrode h -f stage. When a split plate tank coil is employed in the stage in question, the graph of figure 24 should be used. The capacity read from the graph is the total capacity across the tank coil. If the split- stator tuning capacitor is used, each section of the capacitor should have a value of capacity equal to twice the value indicated by the graph. As in the case of figure 23, the values of capacity read on the graph of figure 24 include all residual circuit capacities. For push-pull operation, the correct values of tank circuit capacity may be determined with the aid of figure 25. The capacity values obtained from figure 25 are the effective values across the tank circuit, and if a split- stator tuning capacitor is used, each section of the capacitor should have a value of capacity equal to twice the value indicated by the graph. As in the case of figures 23 and 24, the values of capacity read on the graph of figure 25 include all residual circuit capacities. The tank circuit operates in the same manner whether the tube feeding it is a pentode, beam tetrode, neutralized triode, grounded grid triode, whether it is single ended or push- www.americanradiohistory.com Generation of 262 R -F THE Energy RADIO 20000 Q°6 16000 0 8000 uz á >ú « 6000 0: w J O. a. ú é 1000 2 3 ! 7 10 20 30 30 00 200 300 1000 ) FOR CORRECT VALUES OF TANK CIRCUIT CAPACITANCE C OPERAT NG Q OF 6 WITH PUSH -PULL TANK CIRCUITS Figure 25 PLATE -TANK CIRCUIT ARRANGEMENTS FOR PUSH -PULL STAGES Shown above is recommended tank circuit capacity at operating frequency for a Q of 6. (A) is split-stator tank, each section of which is twice the capacity value read on the graph. (B) is circuit using topped coil for phase reversal. pull, or whether it is shunt fed or series fed. The important thing in establishing the operating Q of the tank circuit is the ratio of the loaded resonant impedance across its terminals to the reactance of the L and the C which make up the tank. Due to the unknowns involved in determining circuit stray capacitances it is sometimes more convenient to determine the value of L required for the proper circuit Q (by the method discussed earlier in this Section) and then to vary the tuned circuit capacitance until resonance is reached. This method is most frequently used in obtaining proper circuit Q in commercial transmitters. The values of Rp for using the charts are easily calculated by dividing the d -c plate supply voltage by the total d -c plate current (expressed in amperes). Correct values of total tuning capacitance are shown in the chart for the different amateur bands. The shunt stray capacitance can be estimated closely enough for all practical purposes. The coil inductance should then be chosen which will produce resonance at the desired frequency with the total calculated tuning capacitance. The Q of a circuit depends upon the resistance in series with the capacitance and inductance. This series resistance is very low for a low -loss coil not loaded by an antenna circuit. The value of Q may be from 100 to 600 under these conditions. Coupling an antenna Effect of Loading on Q circuit has the effect of increasing the series resistance, though in this case the power is consumed as useful radiation by the antenna. Mathematically, the antenna increases the value of R in the expression Q = oiL /R where L is the coil inductance in microhenrys and is the term 2nf, f being in megacycles. The coupling from the final tank circuit to the antenna or antenna transmission line can be varied to obtain values of Q from perhaps 3 at maximum coupling to a value of Q equal to the unloaded Q of the circuit at zero antenna coupling. This value of unloaded Q can be as high as 500 or 600, as mentioned in the preceding paragraph. However, the value of Q = 12 will not be obtained at values of normal d -c plate current in the Class C amplifier stage unless the C -to -L ratio in the tank circuit is correct for that frequency of operation. To determine the required tuning capacitor air gap for a particular amplifier circuit it is first necessary to estimate the peak r-f voltage which will appear between the plates of the tuning capacitor. Then, using figure 26, it is possible to estimate the plate spacing which will be required. The instantaneous r -f voltage in the plate circuit of a Class C amplifier tube varies from nearly zero to nearly twice the d -c plate voltage. If the d -c voltage is being 100 per cent modulated by an audio voltage, the r-f peaks will reach nearly four times the d -c voltage. Tuning Capacitor Air Gap www.americanradiohistory.com HANDBOOK L and Pi Networks RP RA(Q2+1)(txACT) FIGURE 26 USUAL BREAKDOWN RATINGS OF COMMON PLATE SPACINGS Air-gap in Peak voltage inches breakdown .030 1,000 .050 2,000 .070 3,000 .100 4,000 .125 4,500 .150 5,200 .170 6,000 .200 7,500 .250 9,000 .350 11,000 .500 15,000 .700 20,000 Recommended air -gap for use when no d -c voltage appears across plate tank condenser (when plate circuit is shunt fed, or when the plate tank condenser is insulated from ground). D.C. PLATE VOLTAGE 400 600 750 1000 1250 1500 2000 2500 3000 3500 263 C.W. PLATE MOD. .030 .050 .050 .070 .050 .070 .084 .100 .070 RP = Q2 RA (APPROX.) Q=xs__X.. -BL-B.e RA RA XC XL XL =Xc RF +e RP= APPROX. LATE VOLTAG4 'PLATE CURRENT RP= 225 RA = FOR OPERATING Q OF 15 CIRCUIT XC XL= Figure THE * 19 27 NETWORK IMPEDANCE TRANSFORMER The L network is useful with a moderate operating Q for high values of impedance transformation, and it may be used for applications other than in the plate circuit of a tube with relatively low values of operating Q for moderate impedance transformations. Exact and approximate design equations ore given. L .144 .078 .100 .175 .200 .250 .200 .250 .375 .500 .600 should be multiplied by 1.5 for some safety factor when d -c voltage appears Spacings across plate tank condenser. These rules apply to a loaded amplifier or buffer stage. If either is operated without an r -f load, the peak voltages will be greater and can exceed the d-c plate supply voltage. For this reason no amplifier should be operated without load when anywhere near normal d -c plate voltage is applied. If a plate blocking condenser is used, it must be rated to withstand the d -c plate voltage plus any audio voltage. This capacitor should be rated at a d -c working voltage of at least twice the d-c plate supply in a plate modulated amplifier, and at least equal to the d -c supply in any other type of r -f amplifier. between the plate tank circuit of an amplifier and a transmission line, or they may be used to match directly from the plate circuit of an amplifier to the line without the requirement for a tank circuit -provided the network is designed in such a manner that it has sufficient operating Q for accomplishing harmonic attenuation. The L Matching Network L and Pi Matching Networks The L network is of limited utility in impedance matching since its ratio of impedance transformation is fixed at a value equal to (Q2 +1). The operating Q may be relatively low (perhaps 3 to 6) in a matching network between the plate tank circuit of an amplifier and a transmission line; hence impedance transformation ratios of 10 to 1 and even lower may be attained. But when the network also acts as the plate tank circuit of the amplifier stage, as in figure 27, the operating Q should be at least 12 and preferably 15. An operating Q of 15 represents an impedance transformation of 225; this value normally will be too high even for transforming from the 2000 to 10,000 ohm plate impedance of a Class C amplifier stage down to a 50 -ohm transmission The L and pi networks often can be put to advantageous use in accomplishing an impedance match between two differing impedances. Common applications are the matching between a transmission line and an antenna, or between the plate circuit of a single -ended amplifier stage and an antenna transmission line. Such networks may be used to accomplish a match However, the L network is interesting since the basis of design for the pi network. Inspection of figure 27 will show that the L network in reality must be considered as a parallel- resonant tank circuit in which RA represents the coupled -in load resistance; only in this case the load resistance is directly coupled into the tank circuit rather than being inductively coupled as in the conven- 13 -10 line. it forms www.americanradiohistory.com Generation of 264 R -F THE Energy RADIO pacitance may be obtained for an operating Q of 12 by reference to figures 23, 24 and 25. The inductive arm in the pi network can be thought of as consisting of two inductances in series, as illustrated in figure 28. The first is that value of portion of this inductance, inductance which would resonate with C, at the operating frequency -the same as in a conventional tank circuit. However, the actual value of inductance in this arm of the pi network, L10, will be greater than L, for normal values of impedance transformation. For high transformation ratios Lot will be only slightly greater than Li; for a transformation ratio of 1.0, L10t will be twice as great as L. The amount of inductance which must be added to L, to restore resonance and maintain circuit Q is obtained through use of the expression L Roc Ebb = XCZ I Rp XCn -RA Rp RA(Q2+1)-Rp Roc. z Rp = XCi RI12tXG22 XLZ' RAZ ALTOT. XL1+XL2 Q XL, Á for X12 in figure 28. Figure 28 THE PI NETWORK The pi network is valuable for use as an impedance transformer over a wide ratio of transformation values. The operating Q should be at least 12 and preferably 15 to 20 when the circuit is to be used in the plate circuit of Class C amplifier. Design equations are given above. The inductor Ltot represents a single inductance, usually variable, with a value equal to the sum of Lt and L2. a tional arrangement where the load circuit is coupled to the tank circuit by means of a link. When RA is shorted, L and C comprise a conventional parallel- resonant tank circuit, since for proper operation L and C must be resonant in order for the network to present a resistive load to the Class C amplifier. pi impedance matching network, illustrated in figure 28, is much more general in its application than the L network since it offers greater harmonic attenuation, and since it can be used to match a relatively wide range of impedances while still maintaining any desired operating Q. The values of C, and L, in the pi network of figure 28 can be thought of as having the same values of the L network in figure 27 for the same operating Q, but what is more important from the comparison standpoint these values will be the same as in a conventional tank circuit. The value of the capacitance may be determined by calculation, with the operating Q and the load impedance which should be reflected to the plate of the Class C amplifier as the two knowns -or the actual values of the ca- The Pi Network The The peak voltage rating of the main tuning capacitor C, should be the normal value for a Class C amplifier operating at the plate volt- age to be employed. The inductor L101 may be a plug -in coil which is changed for each band of operation, or some sort of variable inductor may be used. A continuously variable slider - type of variable inductor, such as used in certain items of surplus military equipment, may be used to good advantage if available, or a tapped inductor such as used in the ART -13 may be employed. However, to maintain good circuit Q on the higher frequencies when a variable or tapped coil is used on the lower frequencies, the tapped or variable coil should be removed from the circuit and replaced by a smaller coil which has been especially designed for the higher frequency ranges. The peak voltage rating of the output or loading capacitor, C2i is determined by the power level and the impedance to be fed. If a 50 -ohm coaxial line is to be fed from the pi network, receiving -type capacitors will be satisfactory even up to the power level of a plate -modulated kilowatt amplifier. In any event, the peak voltage which will be impressed across the output capacitor is expressed by: Epk2 = 2 R. Wo, where Epk is the peak voltage across the capacitor, R. is the value of resistive load which the network is feeding, and W. is the maximum value of the average power output of the stage. The harmonic attenuation of the pi network is quite good, although an external low -pass filter will be required to obtain harmonic attenuation value upward of 100 db such as normally required. The attenuation to second harmonic energy will be approximately 40 db for an operating Q of 15 for the pi network; the value increases to about 45 db for a 1:1 transformation and falls to about 38 db for an impedance step -down of 80:1, assuming that the operating Q is maintained at 15. www.americanradiohistory.com HANDBOOK Grid Bias 265 RFC CB COAX OUTPUT -B E. Zx[B PLATE LOAD (OHMS) WHERE ES IS PLATE VOLTAGE AND I B IS PLATE CURRENT IN 4MPE II CS Ce- .000252/F MICA CAPACITOR RATED AT TWICE THE D.C. PLATE VOLTAGE RFC i -Ne 28 ENAMELED. CLOSE -WOUND ON I'OIA., RFC2Est mated I. n 1,500 2,000 2,500 3,000 3,500 4,000 4,500 5,000 7 14 21 360 180 90 60 210 105 180 90 120 60 110 56 52 35 45 34 23 30 20 28 65 45 26 33 155 76 38 25 19 135 68 28 280 140 70 47 35 17 15 4.5 2.2 3.5 Mc in puf, 3.5 Mc 7 14 21 28 gal, 6.5 3.2 1.6 3.5 Mc 7 14 21 28 31 8.5 4.2 10.5 5.2 12.5 6.2 14 15.5 7 2.6 3.1 7.8 3.9 2.6 4.5 1.95 2.25 18 9 19 14 20 10 5 6,000 NOTES actuel capacitance setting 45 for C, equals the value in this 23 table minus the published tube 15 output capacitance. Air gap 11 approx. 10 mils 100 v E,,. 90 The 25 Inductance values are for a 12.5 50 -ohm load. For a 70 -ohm 6.2 load, values ore approx. 3% 0.73 0.55 1.08 2.1 1.38 2.05 0.8 1.05 1.7 1.28 3.5 2.3 1.55 1.7 2,400 1,200 600 400 300 2,100 1,060 530 350 265 1,800 900 1,550 760 1,400 700 1,250 630 1,100 560 1,000 500 900 460 450 300 225 380 250 190 350 230 175 320 210 160 280 250 230 700 For 50 -ohm transmission line. 350 Air gap for Cr is approx. 1 175 mil 100 v E,. 185 140 165 125 155 115 120 90 1.800 900 450 300 225 1,500 750 370 250 185 1,300 650 320 215 160 1,100 560 280 1,000 500 250 900 450 220 800 400 200 720 360 180 640 320 160 190 170 125 145 110 130 100 120 110 90 80 500 250 125 85 65 1.1 28 Cr in 2,1 MN, NATIONAL R-100 520 260 130 85 7 14 21 C1 CERAMIC INSULATOR 1,000 µµf, 3.5 Mc in ph, A R-1754 Plate load (ohms) C, 4 -LONG OR NATIONAL 140 3 3.3 2.5 4.1 3.1 higher. For 70 -ohm transmission line. are for a Q of 12. for other values of Q, use Values given are approximations. All components shown in Table L,. Q. C. Q. is higher than 5,000 ohms, it is recommended that the When the estimated plate load and _ 1 Q,, -C,, Q L. components be selected for a circuit Q between 20 and 30. Table 1 Components for Pi- Coupled Final Amplifiers To simplify design pro cedure, a pi- network chart, compiled by M. Seybold, W2RYI (reproduced by courtesy of R.C.A. Tube Division, Harrison, N.J.) is shown in table 1. This chart summarizes the calculations of figure 28 for various values of plate load. Component Chart for Pi- Networks 13 -11 Grid Bias Radio-frequency amplifiers require some form of grid bias for proper operation. Practically all r -f amplifiers operate in such a manner that plate current flows in the forni of short pulses which have a duration of only a fraction of an r -f cycle. To accomplish this with a sinusoidal excitation voltage, the operating grid bias must be at least sufficient to cut off the plate current. In very high efficiency Class C amplifiers the operating bias may be many times the cutoff value. Cutoff bias, it will be recalled, is that value of grid voltage which will reduce the plate current to zero at the plate voltage employed. The method for calculating it has been indicated previously. This theoretical value of cutoff will not reduce the plate current completely to zero, due to the variable-ft tendency or "knee" which is characteristic of all tubes as the cutoff point is approached. Amplitude modulated Class C amplifiers should be operated with the grid bias adjusted to a value greater than twice cutoff at the operating plate voltClass C Bias Generation of 266 FOU R- F THE Energy RADIO FROLIC/RIVER DRIVER Figure 29 GRID -LEAK BIAS The grid leak on an amplifier or multiplier stage may also be used as the shunt feed impedance to the grid of the tube when o high value of grid leak (greater than perhaps 20,000 ohms) is used. When a lower value of grid leak is to be employed, an r -f choke should be used between the grid of the tube and the grid leak to reduce r -f losses in the grid leak resistance. age. This procedure will insure that the tube is operating at a bias greater than cutoff when the plate voltage is doubled on positive modulation peaks. C -w telegraph and FM transmitters can be operated with bias as low as cutoff, if only limited excitation is available and moderate plate efficiency is satisfactory. In a c -w transmitter, the bias supply or resistor should be adjusted to the point which will allow normal grid current to flow for the particular amount of grid driving r -f power available. This form of adjustment will allow more output from the under -excited r -f amplifier than when higher bias is used with corresponding lower values of grid current. In any event, the operating bias should be set at as low a value as will give satisfactory operation, since harmonic generation in a stage increases rapidly as the bias is increased. resistor can be connected Class C amplifier to provide grid -leak bias. This resistor, R, in figure 29, is part of the d -c path Grid -Leak Bias A in the grid circuit of a circuit. The r -f excitation applied to the grid circuit of the tube causes a pulsating direct current to flow through the bias supply lead, due to the rectifying action of the grid, and any current flowing through R, produces a voltage drop across that resistor. The grid of the tube is positive for a short duration of each r -f cycle, and draws electrons from the filament or cathode of the tube during that time. These electrons complete the circuit through the d -c grid return. The voltage drop across the resistance in the grid return provides a negain the grid tive bias for the grid. Grid-leak bias automatically adjusts itself over fairly wide variations of r -f excitation. The value of grid-leak resistance should be such that normal values of grid current will flow at the maximum available amount of r -f Figure 30 COMBINATION GRID -LEAK AND FIXED BIAS Grid -leak bias often is used in conjunction with a fixed minimum value of power supply bias. This arrangement permits the operating bias to be established by the excitation energy, but in the absence of excitation the electrode currents to the tube will be held to safe values by the fixed- minimum power supply bias. If a relatively low value of grid leak is to be used, an r -f choke should be connected between the grid of the tube and the grid leak as discussed in figure 29. excitation. Grid -leak bias cannot be used for grid -modulated or linear amplifiers in which the average d -c grid current is constantly varying with modulation. Safety Bias Grid -leak bias alone provides no protection against e x c e s s i v e plate current in case of failure of the source of r -f grid excitation. A C- battery or C -bias supply can be connected in series with the grid leak, as shown in figure 30. This fixed protective bias will protect the tube in the event of failure of grid excitation. "Zero- bias" tubes do not require this bias source in addition to the grid leak, since their plate current will drop to a safe value when the excitation is removed. resistor can be connected in series with the cathode or center- tapped filament lead of an amplifier to secure automatic bias. The plate current flows through this resistor, then back to the cathode or filament, and the voltage drop across the resistor can be applied to the grid circuit by connecting the grid bias lead to the grounded or power supply end of the resistor R, as shown Cathode Bias A in figure 31. The grounded (B- minus) end of the cathode resistor is negative relative to the cathode by an amount equal to the voltage drop across the resistor. The value of resistance must be so chosen that the sum of the desired grid and plate current flowing through the resistor will bias the tube for proper operation. This type of bias is used more extensively in audio-frequency than in radio -frequency amplifiers. The voltage drop across the resistor www.americanradiohistory.com Protective Circuits HANDBOOK 267 PROM DRIVER Figure Figure 31 R RF STAGE WITH CATHODE BIAS Cathode bias sometimes is advantageous for use it on r -f stage that operates with a relatively small amount of r -f excitation. must be subtracted from the total plate supply voltage when calculating the power input to the amplifier, and this loss of plate voltage in an r-f amplifier may be excessive. A Class A audio amplifier is biased only to approximately one -half cutoff, whereas an r -f amplifier may be biased to twice cutoff, or more, and thus the plate supply voltage loss may be a large percentage of the total available voltage when using low or medium It tubes. Oftentimes just enough cathode bias is employed in an r -f amplifier to act as safety bias to protect the tubes in case of excitation failure, with the rest of the bias coming from a grid leak. Separate Bias Supply An external supply often is used for grid bias, as shown in figure 32. Battery bias gives very good voltage regulation and is satisfactory for grid- modulated or linear amplifiers, which operate at low grid current. In the case of Class C amplifiers which operate with high grid current, battery bias is not satisfactory. This direct current has a charging effect on the dry batteries; after a few months of service the cells will become unstable, bloated, and noisy. A separate a -c operated power supply is commonly used for grid bias. The bleeder resistance across the output of the filter can be made sufficiently low in value that the grid current of the amplifier will not appreciably change the amount of negative grid -bias voltage. Alternately, a voltage regulated grid -bias supply can be used. This type of bias supply is used in Class B audio and Class B r-f linear amplifier service where the voltage regulation in the C-bias supply is important. For a Class C amplifier, regulation is not so important, and an economical design of components in the power supply, therefore, can be utilized. In this case, the bias voltage must be adjusted with normal grid current flowing, as the grid current will raise the bias con- -F 32 STAGE WITH BATTERY BIAS Battery bias is seldom used, due to deterioration of the cells by the reverse grid current. However, it may be used in certain special applications, or the fixed bias voltage may be supplied by a bias power supply. siderably when it is flowing through the bias supply bleeder resistance. 13 -12 Protective Circuits for Tetrode Transmitting Tubes The tetrode transmitting tube requires three operating voltages: grid bias, screen voltage, and plate voltage. The current requirements of these three operating voltages are somewhat interdependent, and a change in potential of one voltage will affect the current drain of the tetrode in respect to the other two voltages. In particular, if the grid excitation voltage is interrupted as by keying action, or if the plate supply is momentarily interrupted, the resulting voltage or current surges in the screen circuit are apt to permanently damage the tube. simple method of obtaining screen voltage is by means of a dropping resistor from the high voltage plate supply, as shown in figure 33. Since the current drawn by the screen is a function of the exciting voltage applied to the tetrode, the screen voltage will rise to equal the plate voltage under conditions of no exciting voltage. If the control grid is overdriven, on the other hand, the screen current may become excessive. In either case, damage to the screen and its associated components may result. In addition, fluctuations in the plate loading of the tetrode stage will cause changes in the screen current of the tube. This will result in screen voltage fluctuations due to the inherently poor voltage regulation of the screen series dropping resistor. These effects become dangerous to tube life if the plate voltage is greater than the screen voltage by a factor of 2 or so. The Series Screen Supply www.americanradiohistory.com A 268 Generation of R -F THE Energy RADIO RFC r NEGATIVE OPERATING Figure 33 8/AS CUTS OFF DROPPING- RESISTOR SCREEN SUPPLY The Clomp Tube CLAMP{ rUBE 4B CLAMP TUBE Figure 34 CLAMP -TUBE SCREEN SUPPLY A clamp tube may be added to the series screen supply, as shown in figure 34. The clamp tube is normally cut off by virtue of the d -c grid bias drop developed across the grid resistor of the tetrode tube. When excitation is removed from the tetrode, no bias appears across the grid resistor, and the clamp tube conducts heavily, dropping the screen voltage to a safe value. When excitation is applied to the tetrode the clamp tube is inoperative, and fluctuations of the plate loading of the tetrode allow the screen voltage to rise to value. Because of this factor, the does not offer complete protection rode. tube could a damaging clamp tube to the tet- A low voltage may be used screen supply of the series screen dropping resistor. This will protect the screen circuit from excessive voltages when the other tetrode operating parameters shift. However, the screen can be easily damaged if plate or bias voltage is removed from the tetrode, as the screen current will reach high values and the screen dissipation will be exceeded. If the screen supply is capable of providing slightly more screen voltage than the tetrode requires for proper operation, a series wattage -limiting resistor may be added to the circuit as shown in figure 35. With this resistor in the circuit it is possible to apply excitation to the tetrode tube with screen voltage present (but in the absence of plate voltage) and still not damage the screen of the tube. The value of the resistor should be chosen so that the product of the voltage applied to the screen of the tetrode times the screen current never exceeds the maximum rated screen dissipation of the tube. The Separate Screen Supply instead piing. The latter is a special form of inductive coupling. The choice of a coupling method depends upon the purpose for which it is to be used. Capacitive coupling between an amplifier or doubler circuit and a preceding driver stage is shown in figure 36. The coupling capacitor, C, isolates the d -c plate supply from the next grid and provides a low impedance path for the r-f energy between the tube being driven and the driver tube. This method of coupling is simple and economical for low power. amplifier or exciter stages, but has certain disadvantages, particularly for high frequency stages. The grid leads in an amplifier should be as short as possible, but this is difficult to attain in the physical arrangement of a high power amplifier with respect to a capacitively- coupled driver stage. Capacitive Coupling One significant disadvantage of capacitive coupling is the difficulty of adjusting the load on the driver stage. Impedance adjustment can be accomplished by tapping the coupling lead a part of the way down on the plate coil of the tuned stage of the driver circuit; but often when this is done Disadvantages of Capacitive Coupling SERIES RESISTOR LOW VOLTAGE SCREEN SUPPLY 13 -13 +B Interstage Coupling Energy is usually coupled from one circuit of a transmitter into another either by capacitive coupling, inductive coupling, or link cou- Figure 35 PROTECTIVE WATTAGE -LIMITING RESISTOR FOR USE WITH LOW- VOLTAGE SCREEN SUPPLY A www.americanradiohistory.com HANDBOOK Figure 36 CAPACITIVE INTERSTAGE COUPLING parasitic oscillation will take place in the stage being driven. One main disadvantage of capacitive coupling lies in the fact that the grid -to- filament capacitance of the driven tube is placed directly across the driver tuned circuit. This condition sometimes makes the r -f amplifier difficult to neutralize, and the increased minimum circuit capacitance makes it difficult to use a reasonable size coil in the v -h -f range. Difficulties from this source can be partially eliminated by using a center-tapped or split stator tank circuit in the plate of the driver stage, and coupling capacitively to the opposite end from the plate. This method places the plate -to- filament capacitance of the driver across one -half of the tank and the grid -tofilament capacitance of the following stage across the other half. This type of coupling is a shown in figure 37. Capacitive coupling can be used to advantage in reducing the total number of tuned circuits in a transmitter so as to conserve space and cost. It also can be used to advantage between stages for driving beam tetrode or pentode amplifier or doubler stages. Inductive coupling (figure 38) results when two coils are electromagnetically coupled to one another. The degree of coupling is controlled by varying the mutual inductance of the two coils, which is accomplished by changing the spacing or the relationship between the axes of the coils. Inductive Coupling Interstage Coupling 269 Figure 37 BALANCED CAPACITIVE COUPLING Balanced capacitive coupling sometimes is useful when it is desirable to use o relatively large inductance in the interstage tank circuit, or where the exciting stage is neutralized as shown above. Inductive coupling is used extensively for coupling r -f amplifiers in radio receivers. However, the mechanical problems involved in adjusting the degree of coupling limit the usefulness of direct inductive coupling in transmitters. Either the primary or the secondary or both coils may be tuned. If the grid tuning capacitor of figure 38 is removed and the coupling increased to the maximum practicable Unity Coupling value by interwinding the turns of the two coils, the circuit insofar as r.f. is concerned acts like that of figure 36, in which one tank serves both as plate tank for the driver and grid tank for the driven stage. The inter-wound grid winding serves simply to isolate the d-c plate voltage of the driver from the grid of the driven stage, and to provide a return for d -c grid current. This type of coupling, illustrated in figure 39, is commonly known as unity coupling. Because of the high mutual inductance, both primary and secondary are resonated by the one tuning capacitor. INTERWOUND Figure 39 "UNITY" INDUCTIVE COUPLING Figure 38 INDUCTIVE INTERSTAGE COUPLING Due to the high value of coupling between the two coils, one tuning capacitor tunes both circuits. This arrangement often is usa ful in coupling from a single -ended to a pushpull stage. 270 Generation of R -F Energy THE LINK COUPLING LINK COUPLING AT AT ..COLD. ENDS. UPPER ENDS "MOT" Figure 40 INTERSTAGE COUPLING BY MEANS OF A used since the two stages may be separated by a considerable distance, since the amount of a coupling between the two stages may he easily varied, and since the capacitances of the two stages may be isolated to permit use of larger inductances in the v-h -f range. special form of inductive coupling which is widely employed in radio transmitter circuits is known as link coupling. A low impedance r -f transmission line couples the two tuned circuits together. Each end of the line is terminated in one or more turns of wire, or links, wound around the coils which are being coupled together. These links should be coupled to each tuned circuit at the point of zero r -f potential, or nodal point. A ground connection to one side of the link usually is used to reduce harmonic coupling, or where capacitive coupling between two circuits must be minimized. Coaxial line is commonly used to transfer energy between the two coupling links, although Twin Lead may be used where harmonic attenuation is not so important. Typical link coupled circuits are shown in figures 40 and 41. Some of the advantages of link coupling are the following: (1) (2) (3) (4) (5) (6) A It eliminates coupling taps on tuned cir- cuits. It permits the use of series power supply connections in both tuned grid and tuned plate circuits, and thereby eliminates the need of shunt -feed r -f chokes. It allows considerable separation between transmitter stages without appreciable r-f losses or stray chassis currents. It reduces capacitive coupling and thereby makes neutralization more easily attainable in r -f amplifiers. It provides semi- automatic impedance matching between plate and grid tuned circuits, with the result that greater grid drive can be obtained in comparison to capacitive coupling. It effectively reduces the coupling of harmonic energy. COLO CENTER ENDS "HOT* Figure "LINK" PUSH -PULL Link interstoge coupling is very commonly Link Coupling RADIO 41 LINK COUPLING The link -coupling line and links can be made of no. 18 push -back wire for coupling between low -power stages. For coupling between higher powered stages the 150 -ohm Twin -Lead transmission line is quite effective and has very low loss. Coaxial transmission is most satisfactory between high powered amplifier stages, and should always be used where harmonic attenuation is important. 13 -14 Radio- Frequency Chokes Radio -frequency chokes are connected in circuits for the purpose of stopping the passage of r -f energy while still permitting a direct current or audio -frequency current to pass. They consist of inductances wound with a large number of turns, either in the form of a solenoid, a series of solenoids, a single universal pie winding, or a series of pie windings. These inductors are designed to have as much inductance and as little distributed or shunt capacitance as possible. The unavoidable small amount of distributed capacitance resonates the inductance, and this frequency normally should be much lower than the frequency at which the transmitter or receiver circuit is operating. R -f chokes for operation on several bands must be designed carefully so that the impedance of the choke will be extremely high (several hundred thousand ohms) in each of the bands. The direct current which flows through the r -f choke largely determines the size of wire to be used in the winding. The inductance of r -f chokes for the v -h -f range is much less than for chokes designed for broadcast and ordinary short -wave operation. A very high inductance r -f choke has more distributed capacitance than a smaller one, with the result www.americanradiohistory.com HANDBOOK Shunt +5G +11V +SG PARALLEL PLATE FEED and Series Feed 271 +Nv -BIAS -BIAS SERIES PLATE FEED PARALLEL BIAS FEED SERIES BIAS FEED Figure 42 ILLUSTRATING PARALLEL AND SERIES PLATE FEED Parallel plate feed is desirable from a safety standpoint since the tank circuit is at ground potential with respect to d.c. However, a Figure 43 ILLUSTRATING SERIES AND PARALLEL BIAS FEED high- impedance r-f choke is required, and the r -t choke must be able to withstand the peak r -f voltage output of the tube. Series plate feed eliminates the requirement for a high -performance r-f choke, but requires the use of a relatively large value of by-pass capacitance at the bottom end of the tank circuit, as contrasted to the moderate value of coupling capacitance which may be used at the top of the tank circuit for parallel plate feed. that it will actually offer less impedance at very high frequencies. Another consideration, just as important as the amount of d.c. the winding will carry, is the r -f voltage which may be placed across the choke without its breaking down. This is a function of insulation, turn spacing, frequency, number and spacing of pies and other factors. Some chokes which are designed to have a high impedance over a very wide range of frequency are, in effect, really two chokes: a u -h -f choke in series with a high -frequency choke. A choke of this type is polarized; that is, it is important that the correct end of the combination choke be connected to the "hot" side of the circuit. Direct-current grid and plate connections are made either by series or parallel leed systems. Simplified forms of each are shown in figures Shunt and Series Feed 42 and 43. Series feed can be defined as that in which the d -c connection is made to the grid or plate circuits at a point of very low r-f potential. Shunt feed always is made to a point of high r -f voltage and always requires a high impedance r-f choke or a relatively high resistance to prevent waste of r -f power. Parallel and 13 -15 Push -Pull Tube Circuits The comparative r -f power output from parallel or push -pull operated amplifiers is the same if proper impedance matching is accomplished, if sufficient grid excitation is available in both cases, and if the frequency of measurement is considerably lower than the frequency limit of the tubes. Operating tubes in parallel has some advantages in transmitters designed for operation below 10 NIc., particularly when tetrode or pentode tubes are to be used. Only one neutralizing capacitor is required for parallel operation of triode tubes, as against two for push -pull. Above about 10 etc., depending upon the tube type, parallel tube operation is not ordinarily recommended with triode tubes. However, parallel operation of grounded -grid stages and stages using low -C beam tetrodes often will give excellent results well into the v -h -f range. Parallel Operation Push -Pull Operation The push -pull connection provides well -balanced circuit insofar as miscellaneous capacitances are concerned; in addition, the circuit can be neutralized more completely, especially in high frequency amplifiers. The L/C ratio in a push pull amplifier can be made higher than in a plate- neutralized parallel -tube operated amplifier. Push -pull amplifiers, when perfectly balanced, have less second-harmonic output than parallel or single -tube amplifiers, but in practice undesired capacitive coupling and circuit unbalance more or less offset the theoretical harmonic-reducing advantages of push pull r -f circuits. a CHAPTER FOURTEEN R -F Feedback Comparatively high gain is required in single sideband equipment because the signal is usually generated at levels of one watt or less. To get from this level to a kilowatt requires about 30 db of gain. High gain tetrodes may be used to obtain this increase with a minimum number of stages and circuits. Each stage contributes some distortion; therefore, it is good practice to keep the number of stages to a minimum. It is generally considered good practice to operate the low level amplifiers below their maximum power capability in order to confine most of the distortion to the last two amplifier stages. R -f feedback can then be utilized to reduce the distortion in the last two stages. This type of feedback is no different from the common audio feedback used in high fidelity sound systems. A sample of the output waveform is applied to the amplifier input to correct the distortion developed in the amplifier. The same advantages can be obtained at radio frequencies that are obtained at audio frequencies when feedback is used. 14 -1 R -F Feedback Circuits R -f feedback circuits have been developed by the Collins Radio Co. for use with linear amplifiers. Tests with large receiving and small transmitting tubes showed that amplifiers using these tubes without feedback developed signal -to- distortion ratios no better than 30 db or so. Tests were run employing cathode follower circuits, such as shown in figure 1A. Lower distortion was achieved, but at the cost of low gain per stage. Since the voltage gain through the tube is less than unity, all gain has to be achieved by voltage step -up in the tank circuits. This gain is limited by the dissipation of the tank coils, since the circuit capacitance across the coils in a typical transmitter is quite high. In addition, the tuning of such a stage is sharp because of the high Q circuits. The cathode follower performance of the tube can be retained by moving the r -f ground B Br Bi>5 `J Figure 1 SIMILAR CATHODE FOLLOWER CIRCUITS HAVING DIFFERENT 272 www.americanradiohistory.com R -F GROUND POINTS. R -F Feedback Circuits 273 B B* BIAS R -F Tuning and loading are accomplished by Cr and C,. C, and L, are tuned in unison to establish the correct degree of feedback. R F OUT BIAS = B4 Figure 4 AMPLIFIER WITH FEEDBACK AND IMPEDANCE MATCHING OUTPUT NETWORK. Figure 2 SINGLE STAGE AMPLIFIER WITH R -F FEEDBACK CIRCUIT B} Figure 3 SINGLE STAGE FEEDBACK AMPLIFIER WITH GROUND RETURN POINT MODIFIED FOR UNBALANCED INPUT AND OUTPUT CONNECTIONS. point of the circuit from the plate to the cathode as shown in figure 1B. Both ends of the input circuit are at high r -f potential so inductive coupling to this type of amplifier is necessary. Inspection of figure 1B shows that by moving the top end of the input tank down on a voltage divider tap across the plate tank circuit, the feedback can be reduced from 100%, as in the case of the cathode follower circuit, down to any desired value. A typical feedback circuit is illustrated in figure 2. This circuit is more practical than those of figure 1, since the losses in the input tank are greatly reduced. A feedback level of 12 db may be achieved as a good compromise between distortion and stage gain. The voltage developed across C= will be three times the grid- cathode voltage. Inductive coupling is required for this circuit, as shown in the illustration. The circuit of figure 3 eliminates the need for inductive coupling by moving the r -f ground to the point common to both tank circuits. The advantages of direct coupling between stages far outweigh the disadvantages of having the r -f feedback voltage appear on the cathode of the amplifier tube. In order to match the amplifier to a load, the circuit of figure 4 may be used. The ratio of XL, to XC, determines the degree of feedback, so it is necessary to tune them in unison when the frequency of operation is changed. Tuning and loading functions are accomplished by varying C2 and G. L5 may also be varied to adjust the loading. Feedback Around o Two -Stage Amplifier The maximum phase shift obtainable over two simple tuned cir- cuits does not exceed 180 degrees, and feedback around a two stage amplifier is possible. The basic circuit of a two stage feedback amplifier is shown in figure 5. This circuit is a conventional two -stage tetrode amplifier except that r -f is fed back from the plate circuit of the PA tube to the cathode of the driver tube. This will reduce the distortion 1 E Figure 5 BASIC CIRCUIT OF TWO -STAGE AMPLIFIER WITH R -F FEEDBACK Feedback voltage is obtained from a voltage divider across the output circuit and applied directly to the cathode of the first tube. The input tank circuit is thus outside the feedback loop. 274 R -F THE RADIO Feedback of both tubes as effectively as using individual feedback loops around each stage, yet will allow a higher level of overall gain. With only two tuned circuits in the feedback loop, it is possible to use 12 to 15 db of feedback and still leave a wide margin for stability. It is possible to reduce the distortion by nearly as many db as are used in feedback. This circuit has two advantages that are lacking in the single stage feedback amplifier. First, the filament of the output stage can now be operated at r -f ground potential. Second, any conventional pi output network may be used. R -f feedback will correct several types of distortion. It will help correct distortion caused by poor power supply regulation, too low grid bias, and limiting on peaks when the plate voltage swing becomes too high. Neutralization The purpose of neutralization of an r -f amplifier stage is to balance out effects of the grid -plate capacitance coupling in the amplifier. In a conventional amplifier using a tetrode tube, the effective input capacity is given by: ond R -F Feedback Input Capacitance where: Ci,, C.. A = Cis + Cy. (1 +A cos e ) = tube input capacitance = grid -plate capacitance = voltage amplification from grid to plate e = phase angle of load In a typical unneutralized tetrode amplifier having a stage gain of 33, the input capacitance of the tube with the plate circuit in resonance is increased by 8 µµfd. due to the unneutralized grid -plate capacitance. This is unimportant in amplifiers where the gain (A ) remains constant but if the tube gain varies, serious detuning and r -f phase shift may result. A grid or screen modulated r -f amplifier is an example of the case where the stage gain varies from a maximum down to zero. The gain of a tetrode r -f amplifier operating below plate current saturation varies with loading so that if it drives a following stage into grid current the loading increases and the gain falls off. The input of the grid circuit is also affected by the grid -plate capacitance, as shown in this equation: Input Resistance - 27rf X C.N ( Asine ) This resistance is in shunt with the grid current loading, grid tank circuit losses, and driving source impedance. When the plate cir- cuit is inductive there is energy transferred from the plate to the grid circuit (positive feedback ) which will introduce negative resistance in the grid circuit. When this shunt negative resistance across the grid circuit is lower than the equivalent positive resistance of the grid loading, circuit losses, and driving source impedance, the amplifier will oscillate. When the plate circuit is in resonance ( phase angle equal to zero) the input resistance due to the grid -plate capacitance becomes infinite. As the plate circuit is tuned to the capacitive side of resonance, the input resistance becomes positive and power is actually transferred from the grid to the plate circuit. This is the reason that the grid current in an unneutralized tetrode r -f amplifier varies from a low value with the plate circuit tuned on the low frequency side of resonance to a high value on the high frequency side of resonance The grid current is proportional to the r -f voltage on the grid which is varying under these conditions. In a tetrode class All amplifier, the effect of grid -plate feedback can be observed by placing a r -f voltmeter across the grid circuit and observing the voltage change as the plate circuit is tuned through resonance. If the amplifier is over -neutralized, the effects reverse so that with the plate circuit tuned to the low frequency side of resonance the grid voltage is high, and on the high frequency side of resonance, it is low. Amplifier Neutralization Check useful "rule of thumb" method of checking neutralization of an amplifier stage (assuming that it is nearly correct to start with) is to tune both grid and plate circuits to resonance. Then, observing the r -f grid current, tune the plate circuit to the high frequency side of resonance. If the grid current rises, more neutralization capacitance is required. Conversely, if the grid current decreases, less capacitance is needed. This indication is very sensitive in a neutralized triode amplifier, and correct neutralization exists when the grid current peaks at the point of plate current dip. In tetrode power amplifiers this indication is less pronounced. Sometimes in a supposedly neutralized tetrode amplifier, there is practically no change in grid voltage as the plate circuit is tuned through resonance, and in some amplifiers it is unchanged on one side of resonance and drops slightly on the other side. Another observation sometimes made is a small dip in the center of a broad peak of grid current. These various effects are probably caused by www.americanradiohistory.com A HANDBOOK R -F R -F Figure 7 NEUTRALIZED AMPLIFIER AND INHERENT FEEDBACK CIRCUIT. Neutralization is achieved by varying the capacity of Cn. - coupling from the plate to the grid circuit through other paths which are not balanced out by the particular neutralizing circuit used. Figure 6 shows an r -f amplifier with negative feed of a One -Stage back. The voltage developed R -F Amplifier across G due to the voltage divider action of G and C, is introduced in series with the voltage developed across the grid tank circuit and is in phase- opposition to it. The feedback can be made any value from zero to 100% by properly choosing the values of C:. and G. For reasons stated previously, it is necessary to neutralize this amplifier, and the relationship for neutralization is: Feedback and Neutralization G == 275 nur Figure 6 SINGLE STAGE R -F AMPLIFIER WITH FEEDBACK RATIO OF DETERMINES C C to C C STAGE NEUTRALIZATION G Feedback Circuits GP G. It is often necessary to add capacitance from plate to grid to satisfy this relationship Figure 7 is identical to figure 6 except that it is redrawn to show the feedback inherent in this neutralization circuit more clearly. G and C replace G and C., and the main plate tank tuning capacitance is G. The circuit of figure 7 presents a problem in coupling to the grid circuit. Inductive coupling is ideal, but the extra tank circuits complicate the tuning of a transmitter which uses several cascaded amplifiers with feedback around each one. The grid could be coupled to a high source impedance such as a tetrode plate, but the driver then cannot use feedback because this would cause the source impedance to be low. A possible solution is to move the circuit ground point from the cathode to the bottom end of the grid tank circuit. The feedback voltage then appears between the cathode and ground ( figure 8 ) . The input can be capacitively coupled, and the plate of the amplifier can be capacitively coupled to the next stage. Also, cathode type transmitting tubes are available that allow the heater to remain at ground po- tential when r -f is impressed upon the cathode. The output voltage available with capacity coupling, of course, is less than the plate cathode r -f voltage developed by the amount of feedback voltage across G. 14 -2 Feedback and Neutralization of Two -Stage R -F a Amplifier Feedback around two r -f stages has the advantage that more of the rube gain can be realized and nearly as much distortion reduction can be obtained using 12 db around two stages as is realized using 12 db around each of two stages separately. Figure 9 shows a basic circuit of a two stage feedback amplifier. Inductive output coupling is used, although a pi- network configuration will also work well. The small feedback voltage required is obtained from the voltage divider C. - G and is applied to the cathode of the driver tube. C. is only a few gcfd., so this feedback voltage divider may be left fixed for a wide frequency range. If the combined tube gain is 160, and 12 db of feedback is desired, the ratio of G to C. is about 40 to 1. This ratio in practice may be 400 µµtd. to 2.5 µµfd., for example. A complication is introduced into this simplified circuit by the cathode -grid capacitance R R-F.N -i Figure 8 UNBALANCED INPUT AND OUTPUT CIRCUITS FOR SINGLE -STAGE R -F AMPLIFIER WITH FEEDBACK www.americanradiohistory.com F our 276 THE RADIO Feedback R -F Figure 9 TWO -STAGE AMPLIFIER WITH FEEDBACK. Included is a capacitor ,C) for neutralizing the cathode -grid capacity of the first tube. by capacitor C and V; is neutralized by the correct ratio of C C-. V. is neutralized , of the first tube which causes an undersired coupling to the input grid circuit. It is necessary to neutralize out this capacitance coupling, as illustrated in figure 9. The relationship for neutralization is: G Cgt G G The input circuit may be made unbalanced by making C. five times the capacity of G. This will tend to reduce the voltage across the coil and to minimize the power dissipated by the coil. For proper balance in this case, G must be five times the grid -filament capacitance of the tube. Except for tubes having extremely small grid -plate capacitance, it is still necessary to properly neutralize both tubes. If the ratio of G to G is chosen to be equal to the ratio of the grid -plate capacitance to the grid -filament capacitance in the second tube (Vg), this tube will be neutralized. Tubes such as a 4X -150A have very low grid -plate capacitance and probably will not need to be neutralized when used in the first (V.) stage. If neutralization is necessary, capacitor G is added for this purpose and the proper value is given by the following relationship: CRO G _ Cet G G C. more feedback from the output stage to overcome. Neutralizing the circuit of figure 9 balances out coupling between the input tank circuit and the output tank circuit, but it does not remove all coupling from the plate Tests For Neutralization circuit to the grid -cathode tube input. This latter coupling is degenerative, so applying a signal to the plate circuit will cause a signal to appear between grid and cathode, even though the stage is neutralized. A bench test for neutralization is to apply a signal to the plate of the tube and detect the presence of a signal in the grid coil by inductive coupling to it. No signal will be present when the stage is neutralized. Of course, a signal could be inductively coupled to the input and neutralization accomplished by adjusting one branch of the neutralizing circuit bridge (G for example) for minimum signal on the plate circuit. Neutralizing the cathode -grid capacitance of the first stage of figure 9 may be accomplished by applying a signal to the cathode of the tube and adjusting the bridge balance for minimum signal on a detector inductively coupled to the input coil. Tuning the two -stage feedback amplifier of figure 9 is accomplished in an unconventional way because the output circuit cannot be tuned for maximum output signal. This is because the output circuit must be tuned so the feedback voltage applied to the cathode is in -phase with the input signal applied to the first grid. When the feedback voltage is not in- phase, the resultTuning o Two -Stage Feedback Amplifier If neither tube requires neutralization, the bottom end of the interstage tank circuit may be returned to r -f ground. The screen and suppressor of the first tube should then be grounded to keep the tank output capacitance directly across this interstage circuit and to avoid common coupling between the feedback on the cathode and the interstage circuit. A slight amount of degeneration occurs in the first stage since the tube also acts as a grounded grid amplifier with the screen as the grounded grid. The p. of the screen is much lower than that of the control grid so that this effect may be unnoticed and would only require slightly ant grid- cathode voltage increases as shown in figure 10. When the output circuit is properly tuned, the resultant grid -cathode voltage on the first tube will be at a minimum, and the voltage on the interstage tuned circuit will also be at a minimum. www.americanradiohistory.com HANDBOOK 1 VOLTAGE VOLTAGE - GRID TO Neutralization 277 CATHODE - INPUT GRID TO GROUND VOLTAGE - CATHODE TO GROUND (PEEDBACM) Figure 12 INTERSTAGE CIRCUIT WITH SEPARATE NEUTRALIZING AND FEEDBACK CIRCUITS. (A. A B Figure 10 VECTOR RELATIONSHIP OF FEEDBACK VOLTAGE Output Circuit Properly Tuned Output Circuit Mis -Tuned The two -stage amplifier may be tuned by placing a r -f voltmeter across the interstage tank circuit ( "hot" side to ground) and tuning the input and interstage circuits for maximum meter reading, and tuning the output circuit for minimum meter reading. If the second tube is driven into the grid current region, the grid current meter may be used in place of the r -f voltmeter. On high powered stages where operation is well into the Class AB region, the plate current dip of the output tube indicates correct output circuit tuning, as in the usual amplifier. Quite often low freq u e n c y parasitics may be found in the interstage circuit of the two -stage feedback amplifier. Oscillation occurs in the first stage due to low frequency feedback in the cathode circuit. R -f chokes, coupling capacitors, and bypass capacitors provide the low frequency tank circuits. When the feedback and second stage neutralizing circuits are combined, it is necessary to use the configuration of figure 11. This circuit has the advantage that only one capacitor (G) is required from the plate of the output tube, thus keeping the added capacitance across the output tank at a minimum. Parasitic Oscillations in the Feedback Amplifier It is convenient, however, to separate these circuits so neutralization and feedback can be adjusted independently. Also, it may be desirable to be able to switch the feedback out of the circuit. For these reasons, the circuit shown in figure 12 is often used. Switch S1 removes the feedback loop when it is closed. A slight tendency for low frequency parasitic oscillations still exists with this circuit. Li should have as little inductance as possible without upsetting the feedback. If the value of Li is too low, it cancels out part of the reactance of feedback capacitor G and causes the feedback to increase at low values of radio frequency. In some cases, a swamping resistor may be necessary across L. The value of this resistor should be high compared to the reactance of G to avoid phase -shift of the r -f feedback. Neutralization 14 -3 Procedure in Feedback -Type Amplifiers Experience with feedback amplifiers has brought out several different methods of neutralizing. An important observation is that when all three neutralizing adjustments are correctly made the peaks and dips of various tuning meters all coincide at the point of circuit resonance. For example, the coincident indications when the various tank circuits are tuned through resonance with feedback operating are: A -When the PA plate circuit is tuned through resonance: -PA plate current dip 2 -Power output peak 3 -PA r -f grid voltage dip 4 -PA grid current dip 1 + BAS Figure 11 INTERSTAGE CIRCUIT COMBINING NEUTRALIZATION AND FEEDBACK NETWORKS. (Note: The PA grid current peaks when feedback circuit is disabled and the tube is heavily driven) 278 R -F R-F THE RADIO Feedback INc Li o O ° cll p FOUT 11(- pi. TY gifIIN 1 c C:7T F Tc. ÌIr ct0 HI, your RFC 1- 1- o T 1 l- 6+ BIAS BIAS Figure 13 TWO -STAGE AMPLIFIER WITH B-When the PA grid circuit is tuned through resonance: Driver plate current dip 2 -PA r -f grid voltage peak 3 -PA grid current peak 4 -PA power output peak 1- C -When the driver grid circuit is tuned through resonance: T ° GF FEED BACK CIRCUIT. 2- Neutralize the grid -plate tance of the driver stage 3- Neutralize the grid -plate capaci- capacitance of the power amplifier (PA) stage 4 -Apply r -f feedback Neutralize driver grid- cathode capacitance 5- 1-Driver r -f grid voltage peak 2-Driver plate current peak These steps will be explained in more detail in the following paragraphs: 3 Step 1. The removal of r -f feedback through the feedback circuit must be complete. The switch ( ) shown in the feedback circuit ( figure 13 ) is one satisfactory method. Since C. is effectively across the PA plate tank circuit it is desirable to keep it across the circuit when feedback is removed to avoid appreciable detuning of the plate tank circuit. Another method that can be used if properly done is to ground the junction of C and C. Grounding this common point through a switch or relay is not good enough because of common coupling through the length of the grounding lead. The grounding method shown in figure 14 is satisfactory. -PA r -f grid current peak -PA plate current peak 5 -PA power output peak 4 Four meters may be employed to measure the most important of these parameters. The meters should be arranged so that the following pairs of readings are displayed on meters located close together for ease of observation of coincident peaks and dips: 2 -PA plate current and power output -PA r -f grid current and PA plate 3 -PA r -f grid 1 current voltage and power output Driver plate current and PA r-f grid voltage 4- The third pair listed above may not be necessary if the PA plate current dip is pronounced. When this instrumentation is provided, the neutralizing procedure is as follows: 1- Remove the r-f feedback FEEDBACK Figure 14 SHORTING DEVICE. Step 2. Plate power and excitation are applied. The driver grid tank is resonated by tuning for a peak in driver r -f grid voltage or driver plate current. The power amplifier grid tank circuit is then resonated and adjusted for a dip in driver plate current. Driver neutralization is now adjusted until the PA r -f grid voltage (or PA grid current) peaks at exactly the point of driver plate current dip. A handy rule for adjusting grid-plate neutralization of a tube without feedback: with all circuits in resonance, detune the plate circuit to the high frequency side of resonance: If grid current to next stage (or power output of the stage under test) increases, more neutralizing capacitance is required and vice versa. If the driver tube operates class A so that a plate current dip cannot be observed, a dif- www.americanradiohistory.com HANDBOOK Neutralization 279 Neutralization The method of neutralization employing a sensitive r -f detector inductively coupled to a tank coil is difficult to apply in some cases because of mechanical construction of the equipment, or because of undesired coupling. Another method for observing neutralization can be used, which appears to be more accurate in actual practice. A sensitive r -f detector such as a receiver is loosely coupled to the grid of the stage being neutralized, as shown in figure 15. The coupling capacitance is of the order of one or two µµfd. It must be small enough to avoid upsetting the neutralization when it is removed because the total grid ground capacitance is one leg of the neutralizing bridge. A signal generator is connected at point S and the receiver at point R. If Coo is not properly adjusted the S -meter on the receiver will either kick up or down as the grid tank circuit is tuned through resonance. Go may be adjusted for minimum deflection of the S-meter as the grid circuit is tuned through Techniques T" Figure 15 FEEDBACK NEUTRALIZING CIRCUIT USING AUXILIARY RECEIVER. ferent neutralizing procedure is necessary This will be discussed in a subsequent section. Step 3. This is the same as step 2 except it is applied to the power amplifier stage. Adjust the neutralization of this stage for a peak in power output at the plate current dip. Step 4. Reverse step back. 1 and apply the r -f feed- resonance. Step 5. Apply plate power and an exciting signal to drive the amplifier to nearly full output. Adjust the feedback neutralization for a peak in amplifier power output at the exact point of minimum amplifier plate current. Decrease the feedback neutralization capacitance if the power output rises when the tank circuit is tuned to the high frequency side of resonance. The above sequence applies when the neutralizing adjustments are approximately correct to start with. If they are far off, some "cut and -try" adjustment may be necessary. Also, the driver stage may break into oscillation if the feedback neutralizing capacitance is not near the correct setting. It is assumed that a single tone test signal for amplifier excitation during the above steps, and that all tank circuits are at resonance except the one being detuned to make the observation. There is some interaction between the driver neutralization and the feedback neutralization so if an appreciable change is made in any adjustment the others should be rechecked. It is important that the grid -plate neutralization be accomplished first when using is used the above procedure, otherwise the feedback neutralization will be off a little, since it partially compensates for that error. The grid-plate capacitance of the tube is then neutralized by connecting the signal generator to the plate of the tube and adjusting Cti of figure 13 for minimum deflection again as the grid tank is tuned through resonance. The power amplifier stage is neutralized in the same manner by connecting a receiver loosely to the grid circuit, and attaching a signal generator to the plate of the tube. The r -f signal can be fed into the amplifier output terminal if desired. Some precautions are necessary when using this neutralization method. First, some driver tubes (the 6CL6, for example) have appreciably more effective input capacitance when in operation and conducting plate current than when in standby condition. This increase in input capacitance may be as great as three or four µµfd, and since this is part of the neutralizing bridge circuit it must be taken into consideration. The result of this change in input capacitance is that the neutralizing adjustment of such tubes must be made when they are conducting normal plate current. Stray coupling must be avoided, and it may prove helpful to remove filament power from the preceding stage or disable its input circuit in some manner. It should be noted that in each of the above adjustments that minimum reaction on the grid is desired, not minimum voltage. Some residual voltage is inherent on the grid when this neutralizing circuit is used. CHAPTER FIFTEEN Amplitude Modulation If the output of a c -w transmitter is varied in amplitude at an audio frequency rate instead of interrupted in accordance with code characters, a tone will be heard on a receiver tuned to the signal. If the audio signal consists of a band of audio frequencies comprising voice or music intelligence, then the voice or music which is superimposed on the radio frequency carrier will be heard on the receiver. When voice, music, video, or other intelligence is superimposed on a radio frequency carrier by means of a corresponding variation in the amplitude of the radio frequency output of a transmitter, amplitude modulation is the result. Telegraph keying of a c -w transmitter is the simplest form of amplitude modulation, while video modulation in a television transmitter represents a highly complex form. Systems for modulating the amplitude of a carrier envelope in accordance with voice, music, or similar types of complicated audio waveforms are many and varied, and will be discussed later on in this chapter. 15-1 Sidebands Modulation is essentially a form of mixing or combining already covered in a previous chapter. To transmit voice at radio frequencies by means of amplitude modulation, the voice frequencies are mixed with a radio frequency carrier so that the voice frequencies are converted to radio frequency sidebands. Though it may be difficult to visualize, the amplitude of the radio frequency carrier does not vary during conventional amplitude modulation. Even though the amplitude of radio frequency voltage representing the composite signal ( resultant of the carrier and sidebands, called the envelope) will vary from zero to twice the unmodulated signal value during full modulation, the amplitude of the carrier component does not vary. Also, so long as the amplitude of the modulating voltage does not vary, the amplitude of the sidebands will remain constant. For this to be apparent, however, it is necessary to measure the amplitude of each component with a highly selective filter. Otherwise, the measured power or voltage will be a resultant of two or more of the components, and the amplitude of the resultant will vary at the modulation rate. If a carrier frequency of 5000 kc. is modulated by a pure tone of 1000 cycles, or 1 kc., two sidebands are formed: one at 5001 kc. (the sum frequency) and one at 4999 kc. (the difference frequency). The frequency of each sideband is independent of the amplitude of the modulating tone, or modulation percentage; the frequency of each sideband is determined only by the frequency of the modulating tone. This assumes, of course, that the transmitter is not modulated in excess of its linear capability. 280 www.americanradiohistory.com Modulation When the modulating signal consists of multiple frequencies, as is the case with voice or music modulation, two sidebands will be formed by each modulating frequency (one on each side of the carrier), and the radiated signal will consist of a band of frequencies. The band width, or channel taken up in the frequency spectrum by a conventional double sideband amplitude-modulated signal, is equal to twice the highest modulating frequency. For example, if the highest modulating frequency is 5000 cycles, then the signal (assuming modulation of complex and varying waveform) will occupy a band extending from 5000 cycles below the carrier to 5000 cycles above the carrier. Frequencies up to at least 2500 cycles, and preferably 3500 cycles, are necessary for good speech intelligibility. If a filter is incorporated in the audio system to cut out all frequencies above approximately 3000 cycles, the band width of a radio- telephone signal can be limited to 6 kc. without a significant loss in intelligibility. However, if harmonic distortion is introduced subsequent to the filter, as would happen in the case of an overloaded modulator or overmodulation of the carrier, new frequencies will be generated and the signal will occupy a band wider than 6 kc. Mechanics of Modulation 15-2 fl 281 f A C.W. OR UNMODULATED CARRIER SINE WAVE AUDIO SIGNAL FROM MODULATOR A 2 ItiÌ% Ì 1TjTI1 1ZIUIÌ 1111111111111111111111111 lA/2 _A /2 III 111 ÌI, 50 t % MODULATED CARRIER A A A 00% MODULATED CARRIER Figure AMPLITUDE MODULATED WAVE Top drawing (Al represents an unmodulated carrier wave; (B) shows the audio output of the modulator. Drawing (C) shows the audio signal impressed on the carrier wave to the extent of 50 per cent modulation; (D) shows the carrier with 100 per cent amplitude modulation. 1 A c -w or unmodulated r-f carrier wave is represented in figure 1A. An audio frequency sine wave is represented by the curve of figure 113. When the two are combined or "mixed," the carrier is said to be amplitude modulated, and a resultant similar to 1C or is obtained. It should be noted that under modulation, each half cycle of r -f voltage differs slightly from the preceding one and the following one; therefore at no time during modulation is the r-f waveform a pure sine wave. This is simply another way of saying that during modulation, the transmitted r -f energy no longer is confined to a single radio frequency. It will be noted that the average amplitude of the peak r -f voltage, or modulation envelope, is the same with or without modulation. This simply means that the modulation is symmetrical (assuming a symmetrical modulating wave) and that for distortionless modulation the upward modulation is limited to a value of twice the unmodulated carrier wave amplitude because the amplitude cannot go below zero on downward portions of the modulation cycle. Figure 1D illustrates the maxi1D obtainable distortionless modulation with a sine modulating wave, the r -f voltage at the peak of the r -f cycle varying from zero to twice the unmodulated value, and the r -f power varying from zero to four times the unmodulated value ( the power varies as the square of the voltage). While the average r -f voltage of the modulated wave over a modulation cycle is the mum same as for the unmodulated carrier, the average power increases with modulation. If the radio frequency power is integrated over the audio cycle, it will be found with 100 per cent sine wave modulation the average r-f power has increased 50 per cent. This additional power is represented by the sidebands, because as previously mentioned, the carrier power does not vary under modulation. Thus, when a 100 -watt carrier is modulated 100 per cent by a sine wave, the total r -f power is 150 watts; 100 watts in the carrier and 25 watts in each of the two sidebands. www.americanradiohistory.com 282 Amplitude Modulation THE long as the relative proporLion of the various sidebands Modulation Percentage RADIO So making up voice modulation is maintained, the signal may be received and detected without distortion. However, the higher the average amplitude of the sidebands, the greater the audio signal produced at the receiver. For this reason it is desirable to increase the modulation percentage, or degree of modulation, to the point where maximum peaks just hit 100 per cent. If the modulation percentage is increased so that the peaks exceed this value, distortion is introduced, and if carried very far, bad interference to signals on nearby channels will result. The amount by which a carrier is being modulated may be expressed either as a modulation factor, varying from zero to 1.0 at maximum modulation, or as a percentage. The percentage of modulation is equal to 100 times the modulation factor. Figure 2A shows a carrier wave modulated by a sine -wave audio tone. A picture such as this might be seen on the screen of a cathode -ray oscilloscope with sawtooth sweep on the horizontal plates and the modulated carrier impressed on the vertical plates. The same carrier without modulation would appear on the oscilloscope screen as ECAR Figure 2 GRAPHICAL DETERMINATION OF MODULATION PERCENTAGE The procedure for determining modulation percentage from the peak voltage points indicated is discussed in the text. Modulation Measurement figure 2B. The percentage of modulation of the positive peaks and the percentage of modulation of the negative peaks can be determined separately from two oscilloscope pictures such as shown. The modulation factor of the positive peaks may be determined by the formula: Emax M = - Ecar Ecar The factor for negative peaks may be determined from this formula: M - Ecar - Emin Ecar In the above two formulas Ern ax is the maximum carrier amplitude with modulation and Ellin is the minimum amplitude; Ecar is the steady -state amplitude of the carrier without modulation. Since the deflection of the spot on a cathode -ray tube is linear with respect to voltage, the relative voltages of these various amplitudes may be determined by measuring the deflections, as viewed on the screen, with a rule calibrated in inches or centimeters. The percentage of modulation of the carrier may be had by multiplying the modulation factor thus obtained by 100. The above procedure assumes that there is no carrier shift, or change in average amplitude, with modulation. If the modulating voltage is symmetrical, such as a sine wave, and modulation is accomplished without the introduction of distortion, then the percentage modulation will be the same for both negative and positive peaks. However, the distribution and phase relationships of harmonics in voice and music waveforms are such that the percentage modulation of the negative modulation peaks may exceed the percentage modulation of the positive peaks, and vice versa. The percentage modulation when referred to without regard to polarity is an indication of the average of the negative and positive peaks. The modulation capability of a transmitter is the maximum percentage to which that transmitter may be modulated before spurious sidebands are generated in the output or before the distortion of the modulating waveform becomes objectionable. The highest modulation capability which any transmitter may have on the negative peaks is 100 per cent. The maximum permissible modulation of many transmitters is less than 100 per cent, especially on positive peaks. The modulation capability of a transmitter may be limited by tubes with inModulation Capability sufficient filament emission, by insufficient excitation or grid bias to a plate- modulated stage, too light loading of any type of amplifier carrying modulated r.f., insufficient power output capability in the modulator, or too much excitation to a grid-modulated stage or a Class B linear amplifier. In any case, the FCC regulations specify that no transmitter be modulated in excess of its modulation capability. Hence, it is desirable to make the modulation capability of a transmitter as near as possible to 100 per cent so that the carrier power may be used most effectively. www.americanradiohistory.com HANDBOOK Speech Waveform Modulation Systems The manner in which the human voice is produced by the vocal cords gives rise to a certain dissymmetry in the waveform of voice sounds when they are picked up by a good -quality microphone. This is especially pronounced in the male voice, and more so on certain voiced sounds than on others. The result of this dissymmetry in the waveform is that the voltage peaks on one side of the average value of the wave will be considerably greater, often two or three times as great, as the voltage excursions on the other side of the zero axis. The average value of voltage on both sides of the wave is, of course, the same. As a result of this dissymmetry in the male voice waveform, there is an optimum polarity of the modulating voltage that must be observed if maximum sideband energy is to be obtained without negative peak clipping and generation of "splatter" on adjacent channels. A double -pole double -throw "phase reversing" switch in the input or output leads of any transformer in the speech amplifier system will permit poling the extended peaks in the direction of maximum modulation capability. The optimum polarity may be determined easily by listening on a selective receiver tuned to a frequency 30 to 50 kc. removed from the desired signal and adjusting the phase reversing switch to the position which gives the least "splatter" when the transmitter is modulated rather heavily. If desired, the switch then may be replaced with permanent wiring, so long as the microphone and speech system are not to be changed. A more conclusive illustration of the lopsidedness of a speech waveform may be obtained by observing the modulated waveform of a radiotelephone transmitter on an oscilloscope. A portion of the carrier energy of the transmitter should be coupled by means of a link directly to the vertical plates of the 'scope, and the horizontal sweep should be a sawtooth or similar wave occurring at a rate of approximately 30 to 70 sweeps per second. With the speech signal from the speech amplifier connected to the transmitter in one polarity it will be noticed that negative -peak clipping -as indicated by bright "spots" in the center of the 'scope pattern whenever the carrier amplitude goes to zero -will occur at a considerably lower level of average modulation than with the speech signal being fed to the transmitter in the other polarity. When the input signal to the transmitter is polarized in such a manner that the "fingers" of the speech wave extend in the direction of positive modulation these fingers usually will be clipped in the plate circuit of the modulator at an acceptable peak modulation level. Dissymmetry 283 The use of the proper polarity of the incoming speech wave in modulating a transmitter can afford an increase of approximately two to one in the amount of speech audio power which may be placed upon the carrier for an amplitude-modulated transmitter for the same amount of sideband splatter. More effective methods for increasing the amount of audio power on the carrier of an AM phone transmitter are discussed later in this chapter. Because the same intelligibility is contained in each of the sidebands associated with a modulated carrier, it is not necessary to transmit sidebands on both sides of the carrier. Also, because the carrier is simply a single radio frequency wave of unvarying amplitude, i t is no t necessary to transmit the carrier if some means is provided for inserting a locally generated carrier at the receiver. When the carrier is suppressed but both upper and lower sidebands are transmitted, it is necessary to insert a locally generated carrier at the receiver of exactly the same frequency and phase as the carrier which was suppressed. For this reason, suppressed carrier double -sideband systems have little practical application. When the carrier is suppressed and only the upper or the lower sideband is transmitted, a highly intelligible signal may be obtained at the receiver even though the locally generated carrier differs a few cycles from the frequency of the carrier which was suppressed at the transmitter. A communications system utilizing but one group of sidebands with carrier suppressed is known as a single sideband system. Such systems are widely used for commercial point to point work, and are being used to an increasing extent in amateur communication. The two chief advantages of the system are: (1) an effective power gain of about 9 db results from putting all the radiated power in intelligence carrying sideband frequencies instead of mostly into radiated carrier, and (2) elimination of the selective fading and distortion that normally occurs in a conventional double - sideband system when the carrier fades and the sidebands do not, or the sidebands fade differently. Single- Sideband Transmission 15 -3 Systems of Amplitude Modulation There are many different systems and methods for amplitude modulating a carrier, but most may be grouped under three general classifications: (1) variable efficiency systems in which the average input to the stage re- www.americanradiohistory.com 284 THE Amplitude Modulation mains constant with and without modulation and the variations in the efficiency of the stage in accordance with the modulating signal accomplish the modulation; (2) constant efficiency systems in which the input to the stage is varied by an external source of modulating energy to accomplish the modulation; and (3) so- called high -efficiency systems in which circuit complexity is increased to obhigh plate circuit efficiency in the modulated stage without the requirement of an external high -level modulator. The various systems under each classification have individual characteristics which make certain ones best suited to particular applications. Since the average input remains constant in a stage employing variable efficiency modulation, and since the average power output of the stage increases with modulation, the additional average power output from the stage with modulation must come from the plate dissipation of the tubes in the stage. Hence, for the best relation between tube cost and power output the tubes employed should have as high a plate dissipation rating per Variable Efficiency Modulation dollar as possible. The plate efficiency in such an ciency in certain types of amplifiers will be as low as 60 per cent, the unmodulated efficiency in such amplifiers will be in the vicinity of 30 per cent. Assuming a typical amplifier having a peak efficiency of 70 per cent, the following figures give an idea of the operation of an idealized efficiency -modulated stage adjusted for 100 per cent sine -wave modulation. It should be kept in mind that the plate voltage is constant at all times, even over the audio cycles. 100 watts 35 watts 35% Input on 100% positive modulation peak (plate current doubles) Efficiency on 200 watts 70% Output tion peak 140 watts 0 watts 100% positive peak on 100% positive modula- Input on 100% negative peak Efficiency on 100% negative peak Output on 100% negative peak 0% 0 watts 100 watts 52.5 watts 52.5% Systems of Efficiency There are many systems of efficiency modulation, but they all have the general limitation discussed in the previous paragraph -so long as the carrier amplitude is to remain constant with and without modulation, the efficiency at carrier level must be not greater than one -half the peak modulation efficiency if the stage is to be capable of 100 per cent modulation. The classic example of efficiency modulation is the Class B linear r -f amplifier, to be discussed below. The other three common forms of efficiency modulation are control grid modulation, screen -grid modulation, and suppressor -grid modulation. In each case, including that of the Class B linear amplifier note that the modulation, or the modulates signal, is impressed on a control electrode of the stage. Modulation amplifier is when going from the unmodulated condition to the peak of the modulation cycle. Hence, the unmodulated efficiency of such an amplifier must always be less than 45 per cent, since the maximum peak efficiency obtainable in a conventional amplifier is in the vicinity of 90 per cent. Since the peak effi- doubled Plate input without modulation Output without modulation Efficiency without modulation Average input with 100% modulation Average output with 100% modulation (35 watts carrier plus 17.5 watts sideband) Average efficiency with 100% modulation RADIO The Class B Linear Amplifier This is the simplest practicable type amplifier for an amplitude -modulated wave or a single -sideband signal. The system possesses the disadvantage that excitation, grid bias, and loading must be carefully controlled to preserve the linearity of the stage. Also, the grid circuit of the tube, in the usual application where grid current is drawn on peaks, presents a widely varying value of load impedance to the source of excitation. Hence it is necessary to include some sort of swamping resistor to reduce the effect of grid- impedance variations with modulation. If such a swamping resistance across the grid tank is not included, or is too high in value, the positive modulation peaks of the incoming modulated signal will tend to be flattened with resultant distortion of the wave being amplified. The Class B linear amplifier has long been used in broadcast transmitters, but recently has received much more general usage in the h -f range for two significant reasons: (a) the Class B linear is an excellent way of increasing the power output of a single -sideband transmitter, since the plate efficiency with full signal will be in the vicinity of 70 per cent, while with no modulation the input to the stage drops to a relatively low value; and (b) the Class B linear amplifier operates with relatively low harmonic output since the grid bias on the stage normally is slightly less www.americanradiohistory.com HANDBOOK Class than the value which will cut off plate current to the stage in the absence of excitation. Since s Class B linear amplifier is biased to extended cutoff with no excitation ( the grid bias at extended cutoff will be approximately equal to the plate voltage divided by the amplification factor for a triode, and will be approximately equal to the screen voltage divided by the grid- screen mu factor for a tetrode or pentode) the plate current will flow essentially in 180 -degree pulses. Due to the relatively large operating angle of plate current flow the theoretical peak plate efficiency is limited to 78.5 per cent, with 65 to 70 per cent representing a range of efficiency normally attainable, and the harmonic output peak output of the r -f envelope should fall to half the value obtained on positive modula- will be low. The carrier power output from a Class B linear amplifier of a normal 100 per cent modulated AM signal will be about one -half the rated plate dissipation of the stage, with optimum operating conditions. The peak output from a Class B linear, which represents the maximum- signal output as a single -sideband amplifier, or peak output with a 100 per cent AM signal, will be about twice the plate dissipation of the tubes in the stage. Thus the carrier -level input power to a Class B linear should be about 1.5 times the rated plate dissipation of the stage. The schematic circuit of a Class B linear amplifier is the same as a conventional singleended or push -pull stage, whether triodes or beam tetrodes are used. However, a swamping resistor, as mentioned before, must be placed across the grid tank of the stage if the operating conditions of the tube are such that appreciable gridcurrent will be drawn on modulation peaks. Also, a fixed source of grid bias must be provided for the stage. A regulated grid -bias power supply is the usual source of negative bias voltage. With grid bias adjusted to the correct value, and with provision for varying the excitation voltage to the stage and the loading of the plate circuit, a fully modulated signal is applied to the grid circuit of the stage. Then with an oscilloscope coupled to the output of the stage, excitation and loading are varied until the stage is drawing the normal plate input and the output wave shape is a good replica of the input signal. The adjustment procedure normally will reAdjustment of a Class 8 Linear Amplifier quire a succession of approximations, until the optimum set of adjustments is attained. Then the modulation being applied to the input signal should be removed to check the linearity. With modulation removed, in the case of a 100 per cent AM signal, the input to the stage should remain constant, and the B Linear Amplifier 285 tion peaks. Class C Grid Modulation . One widely used system of efficiency modulation for communications work is Class C control -grid bias modulation. The distortion is slightly higher than for a properly operated Class B linear amplifier, but the efficiency is also higher, and the distortion can be kept within tolerable limits for communications work. Class C grid modulation requires high plate voltage on the modulated stage, if maximum output is desired. The plate voltage is normally run about 50 per cent higher than for maximum output with plate modulation. The driving power required for operation of a grid -modulated amplifier under these conditions is somewhat more than is required for operation at lower bias and plate voltage, but the increased power output obtainable overbalances the additional excitation requirement. Actually, almost half as much excitation is required as would be needed if the same stage were to be operated as a Class C plate modulated amplifier. The resistor R across the grid tank of the stage serves as swamping to stabilize the r -f driving voltage. At least 50 per cent of the output of the driving stage should be dissipated in this swamping resistor under carrier conditions. A comparatively small amount of audio power will be required to modulate the amplifier stage 100 per cent. An audio amplifier having 20 watts output will be sufficient to modulate an amplifier with one kilowatt input. Proportionately smaller amounts of audio will be required for lower powered stages. However, the audio amplifier that is being used as the grid modulator should, in any case, either employ low plate resistance tubes such as 2A3's, employ degenerative feedback from the output stage to one of the preceding stages of the speech amplifier, or be resistance loaded with a resistor across the secondary of the modulation transformer. This provision of low drive ing impedance in the grid modulator is to insure good regulation in the audio driver for the grid modulated stage. Good regulation of both the audio and the r -f drivers of a grid -modulated stage is quite important if distortion-free modulation approaching 100 per cent is desired, because the grid impedance of the modulated stage varies widely over the audio cycle. A practical circuit for obtaining grid -bias modulation is shown in figure 3. The modulator and bias regulator tube have been combined in a single 6B4G tube. The regulator -modulator tube operates as a cathode - follower. The average d -c voltage www.americanradiohistory.com THE Amplitude Modulation 286 R.F. AMPLIFIER 000 RFC C R ANT a w.w. #IOOA The most satisfactory pro cedure for tuning a stage for grid -bias modulation of the Class C type is as follows. The amplifier should first be neutralized, and any possible tendency toward parasitics under any condition of operation should be eliminated. Then the antenna should be coupled to the plate circuit, the grid bias should be run up to the maximum available value, and the plate voltage and excitation should be applied. The grid bias voltage should then be reduced until the amplifier draws the approximate amount of plate current it is desired to run, and modulation corresponding to about 80 per cent is then applied. If the plate current kicks up when modulation is applied, the grid bias should be reduced; if the plate meter kicks down, increase the MIDGET CHOKE .025 FROM ETC 65J7 47K R2 70 K - T 6UF. it JO 5Y3GT V. 325V. sYQ0OO, 115 SMALL 60-80 MA. V A C B C Figure TRANSFORMER 3 GRID -BIAS MODULATOR per cent. If the antenna coupling is decreased slightly from the condition just described, and the excitation is increased to the point where the amplifier draws the same input, carrier efficiency of 50 per cent is obtainable with tolerable distortion at 90 per cent modulation. Tuning the Grid -Bias Modulated Stage *5 25K IOW AUDIO INPUT RADIO CIRCUIT on the control grid is controlled by the 70, 000 ohm wire -wound potentiometer and this potentiometer adjusts the average grid bias on the modulated stage. However, a -c signal voltage is also impressed on the control -grid of the tube and since the cathode follows this a -c wave the incoming speech wave is superimposed on the average grid bias, thus effecting grid -bias modulation of the r -f amplifier stage. An audio voltage swing is required on the grid of the 6B4G of approximately the same peak value as will be required as bias -voltage swing on the grid -bias modulated stage. This voltage swing will normally be in the region from 50 to 200 peak volts. Up to about 100 volts peak swing can be obtained from a 6SJ7 tube as a conventional speech amplifier stage. The higher voltages may be obtained from a tube such as a 6J5 through an audio transformer of 2:1 or 21/3:1 ratio. With the normal amount of comparatively tight antenna coupling to the modulated stage, a non -modulated carrier efficiency of 40 per cent can be obtained with substantially distortion -free modulation up to practically 100 grid bias. When the amount of bias voltage has been found (by adjusting the fine control, R2, on the bias supply) where the plate meter remains constant with modulation, it is more than probable that the stage will be drawing either too much or too little input. The antenna coupling should then be either increased or decreased (depending on whether the input was too little or too much, respectively) until the input is more nearly the correct value. The bias should then be readjusted until the plate meter remains constant with modulation as before. By slight jockeying back and forth of antenna coupling and grid bias, a point can be reached where the tubes are running at rated plate dissipation, and where the plate milliammeter on the modulated stage remains substantially constant with modulation. The linearity of the stage should then be checked by any of the conventional methods; the trapezoidal pattern method employing a cathode -ray oscilloscope is probably the most satisfactory. The check with the trapezoidal pattern will allow the determination of the proper amount of gain to employ on the speech amplifier. Too much audio power on the grid of the modulated stage should not be used in the tuning -up process, as the plate meter will kick erratically and it will be impossible to make a satisfactory adjustment. Amplitude modulation may be accomplished by varying the screen -grid voltage in a Class amplifier which employs a pentode, beam Screen -Grid Modulation C www.americanradiohistory.com Screen Grid Modulation H A N D B O O K tetrode, or other type of screen -grid tube. The modulation obtained in this way is not especially linear, but screen -grid modulation does offer other advantages and the linearity is quite adequate for communications work. There are two significant and worthwhile advantages of screen -grid modulation for communications work: (1) The excitation requirements for an amplifier which is to be modulated in the screen are not at all critical, and good regulation of the excitation voltage is not required. The normal rated grid- circuit operating conditions specified for Class C c -w operation are quite adequate for screen grid modulation. (2) The audio modulating power requirements for screen -grid modulation are relatively low. A screen -grid modulated r -f amplifier operates as an efficiency -modulated amplifier, the same as does a Class B linear amplifier and a grid -modulated stage. Hence, plate circuit loading is relatively critical as in any efficiency- modulated stage, and must be adjusted to the correct value if normal power output with full modulation capability is to be obtained. As in the case of any efficiency -modulated stage, the operating efficiency at the peak of the modulation cycle will be between 70 and 80 per cent, with efficiency at the carrier level (if the stage is operating in the normal manner with full carrier) about half of the peak- modulation value. There are two main disadvantages of screen grid modulation, and several factors which must be considered if satisfactory operation of the screen -grid modulated stage is to be obtained. The disadvantages are: (I) As mentioned before, the linearity of modulation with respect to screen -grid voltage of such a stage is satisfactory only for communications work, unless carrier- rectified degenerative feed -back is employed around the modulated stage to straighten the linearity of modulation. (2) The impedance of the screen grid to the modulating signal is non -linear. This means that the modulating signal must be obtained from a source of quite low impedance if audio distortion of the signal appearing at the screen grid is to be avoided. Instead of being linear with respect to modulating voltage, as is the plate circuit of a plate modulated Class C amplifier, the screen grid presents approximately a square-law impedance to the modulating signal over the region of signal excursion where the screen is positive with respect to ground. This non -linearity may be explained in the following manner: At the carrier level of a conventional screen modulated stage the plate -voltage swing of the modulated tube is one -half the voltage Screen -Grid Impedance 287 swing at peak- modulation level. This condition must exist in any type of conventional efficiency- modulated stage if 100 per cent positive modulation is to be attainable. Since the plate -voltage swing is at half amplitude, and since the screen voltage is at half its full modulation value, the screen current is relatively low. But at the positive modulation peak the screen voltage is approximately doubled, and the plate -voltage swing also is at twice the carrier amplitude. Due to the increase in plate -voltage swing with increasing screen voltage, the screen current increases more than linearly with increasing screen voltage. In a test made on an amplifier with an 813 tube, the screen current at carrier level was about 6 ma. with screen potential of 190 volts; but under conditions which represented a positive modulation peak the screen current measured 25 ma. at a potential of 400 volts. Thus instead of screen current doubling with twice screen voltage as would be the case if the screen presented a resistive impedance, the screen current became about four times as great with twice the screen voltage. Another factor which must be considered in the design of a screen -modulated stage, if full modulation is to be obtained, is that the power output of a screen -grid stage with zero screen voltage is still relatively large. Hence, if anything approaching full modulation on negative peaks is to be obtained, the screen potential must be made negative with respect to ground on negative modulation peaks. In the usual types of beam tetrode tubes the screen potential must be 20 to 50 volts negative with respect to ground before cut -off of output is obtained. This condition further complicates the problem of obtaining good linearity in the audio modulating voltage for the screen modulated stage, since the screen voltage must be driven negatively with respect to ground over a portion of the cycle. Hence the screen draws no current over a portion of the modulating cycle, and over the major portion of the cycle when the screen does draw current, it presents approximately a square -law impedance. Circuits for ScreenGrid Laboratory analysis of a large number of circuits for accomModulation plishing screen modulation has led to the conclusion that the audio modulating voltage must be obtained from a low- impedance source if low- distortion modulation is to be obtained. Figure 4 shows a group of sketches of the modulation envelope obtained with various types of modulators and also with insufficient antenna coupling. The result of this laboratory work led to the conclusion that the cathode -follower modulator of the basic circuit shown in figure www.americanradiohistory.com 288 Amplitude Modulation THE RADIO ENVELOPE OBTAINED WITH INSUFFICIENT ANTENNA COUPLING +MOO. -50 +5.0. V. APPROX. O Figure 4 SCREEN -MODULATION CIRCUITS Three common screen modulation circuits are illustrated above. All three circuits are capable of giving intelligible voice modulation although the waveform distortion in the circuits of (A) and (B) is likely to be rather severe. The arrangement at (A) is often called "clamp tube" screen modulation; by returning the grid leak on the clomp tube to ground the circuit will give controlled- carrier screen modulation. This circuit has the advantage that it is simple and is well suited to use in mobile transmitters. (B) is an arrangement using a transformer coupled modulator, and offers no particular advantages. The arrangement at (C) is capable of giving good modulation linearity due to the low impedance of the cathode-follower modulator. However, due to the relatively low heater-cathode ratings on tubes suited for use as the modulator, a separate heater supply for the modulator tube normally is required. This limitation makes application of the circuit to the mobile transmitter a special problem, since an isolated heater supply normally is not available. Shown at (D) as an assistance in the tuning of a screen -modulated transmitter (or any efficiency -modulated transmitter for that matter) is the type of modulation envelope which results when loading to the modulated stage is insufficient. is capable of giving good -quality screen grid modulation, and in addition the circuit provides convenient adjustments for the carrier level and the output level on negative modulation peaks. This latter control, P2 in figure 5, allows the amplifier to be adjusted in such a manner that negative -peak clipping cannot take place, yet the negative modulation peaks may be adjusted to a level just above that at which sideband splatter will occur. 5 The Cathode Follower Modulator The cathode follower is ideally suited for use as the modulator for a screen- grid stage since it acts as a relatively low impedance source of modulating voltage for the screen -grid circuit. In addition the cathode follower modulator allows the supply voltage both for the modulator and for the screen grid of the modulated tube to be obtained from the high -voltage supply for the plate of the screen grid tube or beam tetrode. In the usual case the plate supply for the cathode follower, and hence for the screen grid of the modulated tube, may be taken from the bleeder on the high- voltage power supply. A tap on the bleeder may be used, or two resistors may be connected in series to make up the bleeder, with ap- www.americanradiohistory.com HANDBOOK Modulation Systems 289 propriate values such that the voltage applied to the plate of the cathode follower is appropriate for the tube to be modulated. It is important that a bypass capacitor be used from the plate of the cathode - follower modulator to ground. The voltage applied to the plate of the cathode follower should be about 100 volts greater than the rated screen voltage for the tetrode tube as a c -w Class C amplifier. Hence the cathode -follower plate voltage should be about 350 volts for an 815, 2E26, or 829B, about 400 volts for an 807 or 4 -125A, about 500 volts for an 813, and about 600 volts for a 4 -250A or a 4E27. Then potentiometer P1 in figure 5 should be adjusted until the carrier level screen voltage on the modulated stage is about one -half the rated screen voltage specified for the tube as a Class C c -w amplifier. The current taken by the screen of the modulated tube under carrier conditions will be about one - fourth the normal screen current for c -w operation. The only current taken by the cathode follower itself will be that which will flow through the 100,000 -ohm resistor between the cathode of the 6L6 modulator and the negative supply. The current taken from the bleeder on the high -voltage supply will be the carrier level screen current of the tube being modulated (which current passes of course through the cathode follower) plus that current which will pass through the 100,000 -ohm resistor. The loading of the modulated stage should be adjusted until the input to the tube is about 50 per cent greater than the rated plate dissipation of the tube or tubes in the stage. If the carrier -level screen voltage value is correct for linear modulation of the stage, the loading will have to be somewhat greater than that amount of loading which gives maximum output from the stage. The stage may then be modulated by applying an audio signal to the grid of the cathode -follower modulator, while observing the modulated envelope on an oscilloscope. If good output is being obtained, and the modulation envelope appears as shown in figure 4C, all is well, except that P2 in figure 5 should be adjusted until negative modulation peaks, even with excessive modulating signal, do not cause carrier cutoff with its attendant sideband splatter. If the envelope appears as at figure 4D, antenna coupling should be increased while the carrier level is backed down by potentiometer PI in figure 5 until a set of adjustments is obtained which will give a satisfactory modulation envelope as shown in figure 4C. Changing Bands After a satisfactory set of adjustments has been obtained, Figure 5 CATHODE -FOLLOWER SCREEN -MODULATION CIRCUIT A detailed discussion of this circuit, which also is represented in figure 4C, is given in the accompanying text. it is not difficult to readjust the amplifier for operation on different bands. Potentiometers P1 (carrier level), and P2 ( negative peak level) may be left fixed after a satisfactory adjustment, with the aid of the scope, has once been found. Then when changing bands it is only necessary to adjust excitation until the correct value of grid current is obtained, and then to adjust antenna coupling until correct plate current is obtained. Note that the correct plate current for an efficiency -modulated amplifier is only slightly less than the out -of- resonance plate current of the stage. Hence carrier -level screen voltage must be low so that the out -ofresonance plate current will not be too high, and relatively heavy antenna coupling must be used so that the operating plate current will be near the out -of- resonance value, and so that the operating input will be slightly greater than 1.5 times the rated plate dissipation of the tube or tubes in the stage. Since the carrier efficiency of the stage will be only 35 to 40 per cent, the tubes will be operating with plate dissipation of approximately the rated value without modulation. Speech Clipping in The maximum r -f output of an efficiency -modulated stage is limited by the maximum possible plate voltage swing on positive modulation peaks. In the modula lation circuit of figure 5 the minimum output is limited by the minimum voltage which the screen will reach on a negative modulation peak, as set by potentiometer P2 Hence the screen -grid- modulated stage, when using the modulator of figure 5, acts effectively as a speech clipper, provided the modulating signal amplitude is not too much more than that value the Modulated Stage www.americanradiohistory.com 290 THE Amplitude Modulation which will accomplish full modulation. With correct adjustments of the operating conditions of the stage it can be made to clip positive and negative modulation peaks symmetrically. However, the inherent peak clipping ability of the stage should not be relied upon as a means of obtaining a large amount of speech compression, since excessive audio distortion and excessive screen current on the modulated stage will result. Characteristics of Typical Screen Modulated Stage a An important character - istic of the screen-modulated stage, when using the cathode -follower modulator, is that excessive plate voltage on the modulated stage is not required. In fact, full output usually may be obtained with the larger tubes at an operating plate voltage from one half to two- thirds the maximum rated plate voltage for c -w operation. This desirable condition is the natural result of using a low impedance source of modulating signal for the stage. As an example of a typical screen -modulated stage, full output of 75 watts of carrier may be obtained from an 813 tube operating with a plate potential of only 1250 volts. No increase in output from the 813 may be obtained by increasing the plate voltage, since the tube may be operated with full rated plate dissipation of 125 watts, with normal plate efficiency for a screen -modulated stage, 37.5 per cent, at the 1250-volt potential. The operating conditions of a screen -modulated 813 stage are as follows: Plate voltage-1250 volts Plate current -160 ma. Plate input -200 watts Grid current -11 ma. Grid bias -I 10 volts Carrier screen voltage -190 volts Carrier screen current -6 ma. Power output -approx. 75 watts With full 100 per cent modulation the plate current decreases about 2 ma. and the screen current increases about 1 ma.; hence plate, screen, and grid current remain essentially constant with modulation. Referring to figure 5, which was the circuit used as modulator for the 813, (El) measured plus 155 volts, (E2) measured -50 volts, (E3) measured plus 190 volts, (Et) measured plus 500 volts, and the r.m.s. swing at (E5) for full modulation measured 210 volts, which represents a peak swing of about 296 volts. Due to the high positive voltage, and the large audio swing, on the cathode of the 6L6 (triode connected) modulator tube, it is important that the heater of of this tube be fed from a separate filament RADIO transformer or filament winding. Note also that the operating plate -to- cathode voltage on the 6L6 modulator tube does not exceed the 360 volt rating of the tube, since the operating potential of the cathode is considerably above ground potential. Still another form of efficiency modulation may be obtained by applying the audio modulating signal to the suppressor grid of a pentode Class C r -f amplifier. Basically, suppressor -grid modulation operates in the same general manner as other forms of efficiency modulation; carrier plate circuit efficiency is about 35 per cent, and antenna coupling must be rather tight. However, suppressor grid modulation has one sizeable disadvantage, in addition to the fact that pentode tubes are not nearly so widely used as beam tetrodes which of course do not have the suppressor element. This disadvantage is that the screen grid current to a suppressor -grid modulated amplifier is rather high. The high screen current is a natural consequence of the rather high negative bias on the suppressor grid, which reduces the plate- voltage swing and plate current with a resulting increase in the screen current. In tuning a suppressor -grid modulated amplifier, the grid bias, grid current, screen voltage, and plate voltage are about the same as for Class C c -w operation of the stage. But the suppressor grid is biased negatively to a value which reduces the plate- circuit efficiency to about one -half the maximum obtainable from the particular amplifier, with antenna coupling adjusted until the plate input is about 1.5 times the rated plate dissipation of the stage. It is important that the input to the screen grid be measured to make sure that the rated screen dissipation of the tube is not being exceeded. Then the audio signal is applied to the suppressor grid. In the normal application the audio voltage swing on the suppressor will be somewhat greater than the negative bias on the element. Hence suppressor -grid current will flow on modulation peaks, so that the source of audio signal voltage must have good regulation. Tubes suitable for suppressor -grid modulation are: 2E22, 837, 4E27/8001, 5 -125, 804 and 803. A typical suppressor -grid modulated amplifier is illustrated in figure 6. Suppressor -Grid Modulation 15 -4 Input Modulation Systems Constant efficiency variable -input modulation systems operate by virtue of the addition www.americanradiohistory.com HANDBOOK Plate Modulation CARRIER OUTPUT 4E27 '33w R.F INPUT - IG= AIA ISG' 44 M -130 6J5 V. 2.1 STEPUP IP=)OMA. +1500 V. PEAK SWING FOR FULL MODULATION = 210 V. A.F INPUT +300 V -210 Figure V. 6 AMPLIFIER WITH SUPPRESSOR -GRID MODULATION Recommended operating conditions for linear suppressor-grid modulation of a 4E27/ 2578/8001 stage are given on the drawing. of external power to the modulated stage to effect the modulation. There are two general classifications that come under this heading; those systems in which the additional power is supplied as audio frequency energy from a modulator, usually called plate modulation systems, and those systems in which the additional power to effect modulation is supplied as direct current from the plate supply. Under the former classification comes Heising modulation (probably the oldest type of modulation to be applied to a continuous carrier), Class B plate modulation, and series modulation. These types of plate modulation are by far the easiest to get into operation, and they give a very good ratio of power input to the modulated stage to power output; 65 to 80 per cent efficiency is the general rule. It is for these two important reasons that these modulation systems, particularly Class B plate modulation, are at present the most popular for communications work. Modulation systems coming under the second classification are of comparatively recent development but have been widely applied to broadcast work. There are quite a few systems in this class. Two of the more widely used are the Doherty linear amplifier, and the Ter man- Woodyard high- efficiency grid- modulated amplifier. Both systems operate by virtue of a carrier amplifier and a peak amplifier connected together by electrical quarter -wave lines. They will be described later in this section. Plate Modulation Plate modulation is the application of the audio power 291 to the plate circuit of an r -f amplifier. The r -f amplifier must be operated Class C for this type of modulation in order to obtain a radio frequency output which changes in exact accordance with the variation in plate voltage. The r -f ampli fier is 100 per cent modulated when the peak a -c voltage from the modulator is equal to the d.c. voltage applied to the r -f tube. The positive peaks of audio voltage increase the instantaneous plate voltage on the r -f tube to twice the .1c value, and the negative peaks reduce the voltage to zero. The instantaneous plate current to the r -f stage also varies in accordance with the modulating voltage. The peak alternating current in the output of a modulator must be equal to the d -c plate current of the Class C r -f stage at the point of 100 per cent modulation. This combination of change in audio voltage and current can be most easily referred to in terms of audio power in watts. In a sinusoidally modulated wave, the antenna current increases approximately 22 per cent for 100 per cent modulation with a pure tone input; an r -f meter in the antenna circuit indicates this increase in antenna current. The average power of the r -f wave increases 50 per cent for 100 per cent modulation, the efficiency remaining constant. This indicates that in a plate- modulated radiotelephone transmitter, the audio- frequency channel must supply this additional 50 per cent increase in average power for sine -wave modulation. If the power input to the modulated stage is 100 watts, for example, the average power will increase to 150 watts at 100 per cent modulation, and this additional 50 watts of power must be supplied by the modulator when plate modulation is used. The actual antenna power is a constant percentage of the total value of input power. One of the advantages of plate (or power) modulation is the ease with which proper adjustments can be made in the transmitter. Also. there is less plate loss in the r -f amplifier for a given value of carrier power than with other forms of modulation because the plate efficiency is higher. By properly matching the plate impedance of the r -f tube to the output of the modulator, the ratio of voltage and current swing to d -c voltage and current is automatically obtained. The modulator should have a peak voltage output equal to the average d -c plate voltage on the modulated stage. The modulator should also have a peak power output equal to the d -c plate input power to the modulated stage. The average power output of the modulator will depend upon the type of waveform. If the amplifier is being Heising modulated by a Class A stage, the modulator must have an average www.americanradiohistory.com 292 CLASS MODULATED CLASS C R. RADIO THE Amplitude Modulation C AMPLIFIER f. AMPLIFIER CLASS IS MODULATOR +9 Figure 7 HEISING PLATE MODULATION This type of modulation was the first form of plate modulation. It is sometimes known as "constant current" modulation. Because of the effective 1:1 ratio of the coupling choke, it is impossible to obtain 100 per cent modulation unless the plate voltage to the modulated stage is dropped slightly by resistor R. The capacitor C merely byp the audio around R, so that the full a-f output voltage of the modulator is impressed on the Class C stage. power output capability of one -half the input to the Class C stage. If the modulator is a Class B audio amplifier, the average power required of it may vary from one -quarter to more than one -half the Class C input depending upon the waveform. However, the peak power output of any modulator must be equal to the Class C input to be modulated. Heising modulation is the oldest system of plate modulation, and usually consists of a Class A audio amplifier coupled to the r -f amplifier by means of a modulation choke coil, as shown in figure 7. The d.c. plate voltage and plate current in the r-f amplifier must be adjusted to a value which will cause the plate impedance to match Heising Modulation the output of the modulator, since the modulation choke gives a 1 -to -1 coupling ratio. A series resistor, by- passed for audio frequencies by means of a capacitor, must be connected in series with the plate of the r -f amplifier to obtain modulation up to 100 per cent. The peak output voltage of a Class A amplifier does not reach a value equal to the d -c voltage applied to the amplifier and, consequently, the d -c plate voltage impressed across the r -f tube must be reduced to a value equal to MOD. Figure +5 R F. .13 8 PLATE MODULATION This type of modulation is the most flexible in that the loading adjustment can be made in a short period of time and without elaborate test equipment after a change in operating frequency of the Class C amplifier has CLASS B been made. if available a -c peak voltage 100% modulation is to be obtained. A higher degree of distortion can be tolerthe maximum ated in low -power emergency phone transmitters which use a pentode modulator tube, and the series resistor and by -pass capacitor are usually omitted in such transmitters. High -level Class B plate modulation is the least expensive method of plate modulation. Figure 8 shows a conventional Class B plate -modulated Class C amplifier. The statement that the modulator output power must be one -half the Class C input for 100 per cent modulation is correct only if the waveform of the modulating power is a sine wove. Where the modulator waveform is unclipped speech, the average modulator power for 100 per cent modulation is considerably less than one -half the Class C input. Class B Plata Modulation It has been determined experimentally that the ratio of peak to average power in a speech waveform is approximately 4 to 1 as contrasted to a ratio of 2 to 1 in a sine wave. This is due to the high harmonic content of such a waveform, and to the fact that Power Relations in Speech Waveforms www.americanradiohistory.com HANDBOOK Plate Modulation this high harmonic content manifests itself by making the wave unsymmetrical and causing sharp peaks or "fingers" of high energy content to appear. Thus for unclipped speech, the average modulator plate current, plate dissipation, and power output are approximately one -half the sine wave values for a given peak output power. Both peak power and average power are necessarily associated with waveform. Peak power is just what the name implies; the power at the peak of a wave. Peak power, although of the utmost importance in modulation, is of no great significance in a -c power work, except insofar as the average power may be determined from the peak value of a known wave form. There is no time element implied in the definition of peak power; peak power may be instantaneous -and for this reason average power, which is definitely associated with time, is the important factor in plate dissipation. It is possible that the peak power of a given waveform be several times the average value; for a sine wave, the peak power is twice the average value, and for unclipped speech the peak power is approximately four times the average value. For 100 per cent modulation, the peak (instantaneous) audio power must equal the Class C input, although the average power for this value of peak varies widely depending upon the modulator waveform, being greater than 50 per cent for speech that has been clipped and filtered, 50 per cent for a sine wave, and about 25 per cent for typical unclipped speech tones. Modulation Transformer Calculations The modulation transformer is a device for matching the load impedance of the Class C amplifier to the recommended load impedance of the Class B modulator tubes. Modulation transformers intended for communications work are usually designed to carry the Class C plate current through their secondary windings, as shown in figure 8. The manufacturer's ratings should be consulted to insure that the d-c plate current passed through the secondary winding does not exceed the maximum rating. A detailed discussion of the method of making modulation transformer calculations has been given in Chapter Six. However, to emphasize the method of making the calculation, an additional example will be given. Suppose we take the case of a Class C amplifier operating at a plate voltage of 2000 with 225 ma. of plate current. This amplifier would present a load resistance of 2000 divided by 0.225 amperes or 8888 ohms. The plate power input would be 2000 times 0.225 or 450 watts. By reference to Chapter Six we see that 293 a pair of 811 tubes operating at 1500 plate volts will deliver 225 watts of audio output. The plate -to -plate load resistance for these tubes under the specified operating conditions is 18,000 ohms. Hence our problem is to match the Class C amplifier load resistance of 8888 ohms to the 18,000 -ohm load resistance required by the modulator tubes. A 200 -to -300 watt modulation transformer will be required for the job. If the taps on the transformer are given in terms of impedances it will only be necessary to connect the secondary for 8888 ohms (or a value approximately equal to this such as 9000 ohms) and the primary for 18,000 ohms. If it is necessary to determine the proper turns ratio required of the transformer it can be determined in the following manner. The square root of the impedance ratio is equal to the turns ratio, hence: 8888 18000 = V 0.494 = 0.703 The transformer must have a turns ratio of approximately 1- to -0.7 step down, total primary to total secondary. The greater number of turns always goes with the higher impedance, and vice versa. Plate- andScreen Modulation only the plate of a screen -grid tube is modulated, it is impossible to obtain high -percentage linear modulation under ordinary conditions. The plate current of such a stage is not linear with plate voltage. However, if the screen is modulated simultaneously with the plate, the instantaneous screen voltage drops in proportion to the drop in the plate voltage, and linear modulation can then be obtained. Four satisfactory circuits for accomplishing combined plate and screen modulaWhen tion are shown in figure 9. The screen r -f by -pass capacitor C2 should not have a greater value than 0.005 µfd., preferably not larger than 0.001 tad. It should be large enough to bypass effectively all r -f voltage without short- circuiting high- frequency audio voltages. The plate by -pass capacitor can be of any value from 0.002 µfd. to 0.005 µfd. The screen -dropping resistor, 111. should reduce the applied high voltage to the value specified for operating the particular tube in the circuit. Capacitor C1 is seldom required yet some tubes may require this capacitor in order to keep C2 from attenuating the high frequencies. Different values between .0002 and .002 µfd. should be tried for best results. Figure 9C shows another method which uses a third winding on the modulation transformer, through which the screen -grid is connected to www.americanradiohistory.com 294 RADIO THE Amplitude Modulation E 3 3 B+ S.G. Figure B+ 9 PLATE MODULATION OF A BEAM TETRODE OR SCREEN -GRID TUBE These alternative arrangements for plate modulation of tetrodes or pentodes are discussed in detail in the text. The arrangements shown at (B) or (D) are recommended for most applications. a low- voltage power supply. The ratio of turns between the two output windings depends upon the type of screen -grid tube which is being modulated. Normally it will be such that the screen voltage is being modulated 60 per cent when the plate voltage is receiving 100 per cent modulation. If the screen voltage is derived from a dropping resistor ( not a divider) that is bypassed for r.f. but not a.f., it is possible to secure quite good modulation by applying modulation only to the plate. Under these conditions, the screen tends to modulate itself, the screen voltage varying over the audio cycle as a result of the screen impedance increasing with plate voltage, and decreasing with a decrease in plate voltage. This circuit arrangement is illustrated in figure 9B. A similar application of this principle is shown in figure 9D. In this case the screen voltage is fed directly from a low- voltage supply of the proper potential through a choke L. A conventional filter choke having an inductance from 10 to 20 henries will be satisfactory for L. To afford protection of the tube when plate voltage is not applied but screen voltage is supplied from the exciter power supply, when using the arrangement of figure 9D, a resistor of 3000 to 10,000 ohms can be connected in series with the choke L. In this case the screen supply voltage should be at least 1%Z times as www.americanradiohistory.com HANDBOOK Cathode Modulation much as is required tor actual screen voltage, and the value of resistor is chosen such that with normal screen current the drop through the resistor and choke will be such that normal screen voltage will be applied to the tube. When the plate voltage is removed the screen current will increase greatly and the drop through resistor R will increase to such a value that the screen voltage will be lowered to the point where the screen dissipation on the tube will not be exceeded. However, the supply voltage and value of resistor R must be chosen carefully so that the maximum rated screen dissipation cannot be exceeded. The maximum possible screen dissipation using this arrangement is equal to: W = E' /4R where E is the screen supply voltage and R is the combined resistance of the resistor in figure 9D and the d -c resistance of the choke L. It is wise, when using this arrangement to check, using the above formula, to see that the value of W' obtained is less than the maximum rated screen dissipation of the tube or tubes used in the modulated stage. This same system can of course also be used in figuring the screen supply circuit of a pentode or tetrode amplifier stage where modulation is not to be applied. The modulation transformer for plate -andscreen- modulation, when utilizing a dropping resistor as shown in figure 9A, is similar to the type of transformer used for any plate modulated phone. The combined screen and plate current is divided into the plate voltage in order to obtain the Class C amplifier load impedance. The peak audio power required to obtain 100 per cent modulation is equal to the d-c power input to the screen, screen resistor, and plate of the modulated r -f stage. 15 -5 Cathode Modulation Cathode modulation offers a workable compromise between the good plate efficiency but expensive modulator of high -level plate modulation, and the poor plate efficiency but inexpensive modulator of grid modulation. Cathode modulation consists essentially of an admixture of the two. The efficiency of the average well- designed plate -modulated transmitter is in the vicinity of 75 to 80 per cent, with a compromise perhaps at 77.5 per cent. On the other hand, the efficiency of a good grid-modulated transmitter may run from 28 to maybe 40 per cent, with the average falling at about 34 per cent. Now since cathode modulation consists of simultaneous grid and plate modulation, in phase with each other, we can theoretically obtain any efficiency from about 34 to 77.5 per cent from our cathode -modulated stage, depending 295 upon the relative percentages of grid and plate modulation. Since the system is a compromise between the two fundamental modulation arrangements, a value of efficiency approximately half way between the two would seem to be the best compromise. Experience has proved this to be the case. A compromise efficiency of about 56.5 per cent, roughly half way between the two limits, has proved to be optimum. Calculation has shown that this value of efficiency can be obtained from a cathode -modulated amplifier when the audio- frequency modulating power is approximately 20 per cent of the d-c input to the cathode -modulated stage. Series cathode modulation is ideally suited as an economiModulator cal modulating arrangement for a high -power triode c -w transmitter. The modulator can be constructed quite compactly and for a minimum component cost since no power supply is required for it. When it is desired to change over from c -w to 'phone, it is only necessary to cut the series modulator into the cathode return circuit of the c -w amplifier stage. The plate voltage for the modulator tubes and for the speech amplifier is taken from the cathode voltage drop of the modulated stage across the modulator unit. Figure 10 shows the circuit of such a modulator, designed to cathode modulate a Class C amplifier using push -pull 810 tubes, running at a supply voltage of 2500, and with a plate input of 660 watts. The modulated stage runs at about 50% efficiency, giving a power output of nearly 350 watts, fully modulated. The voltage drop across the cathode modulator is 400 volts, allowing a net plate to cathode voltage of 2100 volts on the final amplifier. The plate current of the 810's should be about 330 ma., and the grid current should be approximately 40 ma., making the total cathode current of the modulated stage 370 ma. Four parallel 6L6 modulator tubes can pass this amount of plate current without difficulty. It must be remembered that the voltage drop across the cathode modulator is also the cathode bias of the modulated stage. In most cases, no extra grid bias is necessary. If a bias supply is used for c -w operation, it may be removed for cathode modulation, as shown in figure 11. With low-mu triodes, some extra grid bias (over and above that amount supplied by the cathode modulator) may be needed to achieve proper linearity of the modulated stage. In any case, proper operation of a cathode modulated stage should be determined by examining the modulated output waveform of the stage on an oscilloscope. An Economical Series Cathode Excitation r-f driver for a cathode -modulated stage should have about The www.americanradiohistory.com 296 THE Amplitude Modulation RADIO TO CATHODE MODULATED STAGE 6L6 6L6 6AU6 6AU6 6 6 L6 L6 500K .002 T 10 K l W °T CAUTION ALL RESISTORS 0.5 wArr (INCEST OTHERWISE NOTED ALL CAPACITORS IN LIP UNLESS OTHERWISE NOTED. Figure - FILAMENTS OF OL° rueES MUST SE Al OPERATING TEMPERATURE BEFORE PLATE VOLTAGE IS APPLIED TO MODULATED AMPLIFIER. 10 SERIES CATHODE MODULATOR FOR A HIGH -POWERED TRIODE R -F AMPLIFIER the same power output capabilities as would be required to drive a c -w amplifier to the same input as it is desired to drive the cathode modulated stage. However, some form of excitation control should be available since the amount of excitation power has a direct bearing on the linearity of a cathode -modulated amplifier stage. If link coupling is used between the driver and the modulated stage, variation in the amount of link coupling will afford apple excitation variation. If much less than 40% plate modulation is employed, the stage begins to resemble a grid -bias modulated stage, and the necessity for good r -f regulation will apply. Cathode modulation has not proved too satisfactory for use with beam tetrode tubes. This is a result of the small excitation and grid swing requirements for such tubes, plus the fact that some means for holding the screen voltage at the potential of the cathode as far as audio is concerned is usually necessary. Because of these factors, cathode modulation is not recommended for use with tetrode r -f amplifiers. Cathode Modulation of Tetrodes 15 -6 The Doherty and the Terman- Woodyard Modulated Amplifiers These two amplifiers will be described together since they operate upon very similar principles. Figure 12 shows a greatly simplified schematic diagram of the operation of both types. Both systems operate by virtue of a carrier tube (V, in both figures 12 and 13) which supplies the unmodulated carrier, and whose output is reduced to supply negative peaks, and a peak tube (V2) whose function is to supply approximately half the positive peak of the modulation cycle and whose additional function is to lower the load impedance on the carrier tube so that it will be able to supply the other half of the positive peak of the modulation cycle. The peak tube is enabled to increase the output of the carrier tube by virtue of an impedance inverting line between the plate circuits of the two tubes. This line is designed to have a characteristic impedance of one -half the value of load into which the carrier tube operates under the carrier conditions. Then a load of one -half the characteristic impedance of the quarter -wave line is coupled into the output. By experience with quarter -wave lines in antenna -matching circuits we know that such a line will vary the impedance at one end of the line in such a manner that the geometric mean between the two terminal impedances will be equal to the characteristic impedance of the line. Thus, if we have a value of load of one -half the characteristic impedance of the line at one end, the other end of the line will present a value of Juice the characteristic impedance of the lines to the carrier tube V,. This is the situation that exists under the carrier conditions when the peak tube merely floats across the load end of the line and contributes no power. Then as a positive peak of modulation comes along, the peak tube starts to contribute power to the load until at the peak of the modulation cycle it is contributing enough power so that the impedance at the load end of the line is equal to R, instead of www.americanradiohistory.com HANDBOOK R Doherty Amplifier F. AMPLIFIER vi 297 ELECTRICAL 5/4 (LINE ZO'R 040 LOAD BIAS SUPPLY FOR C. W. MIC BAIA! BAUE PHONE Figure 12 DIAGRAMMATIC REPRESENTATION OF THE DOHERTY LINEAR PHONE -ELE'S CATHODE MODULATOR Figure 11 MODULATOR INSTALLATION SHOWING PHONE -C.W. TRANSFER SWITCH CATHODE desirable phase shift of 90° between the plate circuits of the carrier and peak tubes, an equal and opposite phase shift must be introduced in the exciting voltage to the grid circuits of the two tubes so that the resultant output in the plate circuit will be in phase. This additional phase shift has been indicated in figure 12 and a method of obtaining it has been shown in figure 13. The difference between the Doherty linear amplifier and the TermanGrid Modulator Woodyard grid -modulated amplifier is the same as the difference between any linear and grid -modulated stages.Modulated r.f.is applied to the grid circuit of the Doherty linear amplifier with the carrier tube biased to cutoff and the peak tube biased to the point where it draws substantially zero plate current at the carrier condition. Comparison Between Linear and the R/2 that is presented under the carrier conditions. This is true because at a positive modulation peak (since it is delivering full power) the peak tube subtracts a negative resistance of R/2 from the load end of the line. Now, since under the peak condition of modulation the load end of the line is terminated in R ohms instead of R /2, the impedance at the carrier -tube will be reduced from 2R ohms to R ohms. This again is due to the impedance inverting action of the line. Since the load resistance on the carrier tube has been reduced to half the carrier value, its output at the peak of the modulation cycle will be doubled. Thus we have the necessary condition for a 100 per cent modulation peak; the amplifier will deliver four times as much power as it does under the carrier conditions. On negative modulation peaks the peak tube does not contribute; the output of the carrier tube is reduced until on a 100 per cent negative peak its output is zero. While an electrical quarter wave line (consisting of a pi Line network with the inductance and capacitance units having a reactance equal to the characteristic impedance of the line) does have the desired impedance- inverting effect, it also has the undesirable effect of introducing a 90° phase shift across such a line. If the shunt elements are capacitances, the phase shift across the line lags by 90 °; if they are inductances, the phase shift leads by 90 °. Since there is an unThe Electrical Quorter -Wave In the Terman -Woodyard grid-modulated amthe carrier tube runs Class C with comparatively high bias and high plate efficiency, while the peak tube again is biased so that it draws almost no plate current. Unmodulated r.f. is applied to the grid circuits of the two tubes and the modulating voltage is inserted in series with the fixed bias voltages. From one -half to two -thirds as much audio voltage is required at the grid of the peak tube as is required at the grid of the carrier tube. plifier The resting carrier efficiency of the grid- modulated amplifier may run as high as is obtainable in any Class C stage, 80 per cent or better. The resting carrier efficiency of the linear will be about as good as is obtainable in any Class 13 amplifier, 60 to 70 per cent. The overall efficiency of the bias -modulated amplifier at 100 per cent modulation will run about 75 per cent; of the linear, about 60 per cent. In figure 13 the plate tank circuits are detuned enough to give an effect equivalent to the shunt elements of the quarter -wave "line" of figure 12. At resonance, the coils L, and L2 in the grid circuits of the two tubes have Operating Efficiencies www.americanradiohistory.com 298 THE Amplitude Modulation NC Q LI ó EXCITATION BIAS r a L30 d V.I-1_ other undesirable features which make their use impracticable alongside the more conventional modulation systems. Nearly all these circuits have been published in the 1.R.E. Proceedings and the interested reader can refer to them in back copies of that journal. Ci 15 -7 TO O T ANT. Tc3 2 Figure 13 SIMPLIFIED SCHEMATIC OF A "HIGH EFFICIENCY" AMPLIFIER The basic system, comprising a "carrier" tube and a "peak" tube interconnected by lumped -constant quarter -wave lines, is the some for either grid-bias modulation or for use as a linear amplifier of a modulated wave. each an inductive reactance equal to the capacitive reactance of the capacitor C1, Thus we have the effect of a pi network consisting of shunt inductances and series capacitance. In the plate circuit we want a phase shift of the same magnitude but in the opposite direction; so our series element is the inductance L3 whose reactance is equal to the characteristic impedance desired of the network. Then the plate tank capacitors of the two tubes C2 and C3 are increased an amount past resonance, so that they have a capacitive reactance equal to the inductive reactance of the coil L3. It is quite important that there be no coupling between the inductors. Although both these types of amplifiers are highly efficient and require no high -level audio equipment, they are difficult to adjust- particularly so on the higher frequencies -and it would be an extremely difficult problem to design a multiband transmitter employing the circuit. However, the grid -bias modulation system has advantages for the high -power transmitter which will be operated on a single fre- quency band. Other High- Efficiency Modulation Systems Many other high- efficiency modulation systems have been described since about 1936. The majority of these, however have received little application either by commercial interests or by amateurs. In most cases the circuits are difficult to adjust, or they have RADIO Speech Clipping Speech waveforms are characterized by frequently recurring high -intensity peaks of very short duration. These peaks will cause over modulation if the average level of modulation on loud syllables exceeds approximately 30 per cent. Careful checking into the nature of speech sounds has revealed that these high intensity peaks are due primarily to the vowel sounds. Further research has revealed that the vowel sounds add little to intelligibility, the major contribution to intelligibility coming from the consonant sounds such as v, b, k, s, t, and 1. Measurements have shown that the power contained in these consonant sounds may be down 30 db or more from the energy in the vowel sounds in the same speech passage. Obviously, then, if we can increase the relative energy content of the consonant sounds with respect to the vowel sounds it will be possible to understand a signal modulated with such a waveform in the presence of a much higher level of background noise and interference. Experiment has shown that it is possible to accomplish this desirable result simply by cutting off or clipping the high- intensity peaks and thus building up in a relative manner the effective level of the weaker sounds. Such clipping theotetically can be accomplished simply by increasing the gain of the speech amplifier until the average level of modulation on loud syllables approaches 90 per cent. This is equivalent to increasing the speech power of the consonant sounds by about 10 times or, conversely, we can say that 10 db of clipping has been applied to the voice wave. However, the clipping when accomplished in this manner will produce higher order side bands known as "splatter," and the transmitted signal would occupy a relatively tremendous slice of spectrum. So another method of accomplishing the desirable effects of clipping must be employed. A considerable reduction in the amount of splatter caused by a moderate increase in the gain of the speech amplifier can be obtained by poling the signal from the speech amplifier to the transmitter such that the high- intensity peaks occur on upward or positive modulation. Overloading on positive modulation peaks produces less splatter than the negative -peak clipping which occurs with overloading on the www.americanradiohistory.com Speech HANDBOOK Clipping 299 Figure 14 SPEECH -WAVEFORM AMPLITUDE MODULATION Showing the effect of using the proper polarity of a speech wave for modulating a transmitter. (A) shows the effect of proper speech polarity on a transmitter having an upward modulation capability of greater than 100 per cent. (B) shows the effect of using proper speech polarity on a transmitter having an upward modulation capability of only 100 per cent. Both these conditions will give a clean signal without objectionable splatter. (C) shows the effect of the use of improper speech polarity. This condition will cause serious splatter due to negative -peak clipping in the modulated amplifier stage. 1001b NEG MODULATION _100 Q AVERAGE LEVEL 100 % NEG. MODULATION 100 % POS. MODULATION AVERAGE LEVEL r NEGATIVE PEAR CLIPPING negative peaks of modulation. This aspect of the problem has been discussed in more detail in the section on Speech Waveform Dissymmetry earlier in this chapter. The effect of feeding the proper speech polarity from the speech amplifier is shown in figure 14. A much more desirable and effective method of obtaining speech clipping is actually to employ a clipper circuit in the earlier stages of the speech amplifier, and then to filter out the objectionable distortion components by means of a sharp low -pass filter having a cut-off frequency of approximately 3000 cycles. Tests on clipper -filter speech systems have shown that 6 db of clipping on voice is just noticeable, 12 db of clipping is quite acceptable, and values of clipping from 20 to 25 db are tolerable under such conditions that a high degree of clipping is necessary to get through heavy QRM or QRN. A signal with 12 db of clipping doesn't sound quite natural but it is not unpleasant to listen to and is much more readable than an unclipped signal in the presence of strong interference. The use of a clipper- filter in the speech amplifier, to be completely effective, requires that phase shift between the clipper- filter stage and the final modulated amplifier be kept %b POS. MODULAT I 100 %b NEG. MODULATION 1 a minimum. However, if there is phase shift after the clipper- filter the system does not completely break down. The presence of phase shift merely requires that the audio gain following the clipper- filter be reduced to the point where the cant applied to the clipped speech waves still cannot cause overmodulation. This effect is illustrated in figures 15 and 16. The cant appearing on the tops of the square waves leaving the clipper -filter centers about the clipping level. Hence, as the frequency being passed through the system is lowered, the amount by which the peak of the canted wave exceeds the clipping level is increased. to In a normal transmitter having a moderate amount of phase shift the cant applied to the tops of the waves will cause overmodulation on frequencies below those for which the gain following the clipper -filter has been adjusted unless remedial steps have been taken. The following steps are advised: (1) Introduce bass suppression into the speech amplifier ahead of the clipper- filter. Phase Shift Correction (2) improve the low- frequency response characteristic insofar as it is possible in the www.americanradiohistory.com 300 THE Amplitude Modulation RADIO POSITIVE CLIPPING LEVEL AVERAGE LEVEL IttGATIVE ÇLIPPMGJ-41F,L INCOMING SPEECH WAVE POSITIVE CLIPPING LEVEL AVERAGE LEVEL NEGATIVE CLIPPING LEVEL CLIPPED AND FILTERED SPEECH WAVE _100% POSITIVE MODULATION 70% POSITIVEMOOUÇATIQN AVERAGE LEVEL Figure 15 ACTION OF A CLIPPER -FILTER ON A SPEECH WAVE The drawing (A) shows the incoming speech wave before it reaches the clipper stage. (B) shows the output of the clipper- filter, illustrating the manner in which the peaks are clipped and then the sharp edges of the clipped wave removed by the filter. (C) shows the effect of p hase shift In the stages following the clipper- filter. (C) also shows the manner in which the transmitter may be adjusted for 100 per cent modulation of' the "canted" peaks of the wave, the sloping top of the wave reaching about 70 per cent modulation. 70 % NEGATIVE MODULATION 100% NEGATIVE MODULATION MODULATED WAVE AFTER UNDERGOING PHASE SHIFT stages following the clipper -filter. Feeding the plate current to the final amplifier through a choke rather than through the secondary of the modulation transformer will help materially. Even with the normal amount of improvement which can be attained through the steps mentioned above there will still be an amount of wave cant which must be compensated in some manner. This compensation can be done in either of two ways. The first and simpler way is as follows: (1) Adjust the speech gain ahead of the clipper- filter until with normal talking into the microphone the distortion being introduced by the clipper -filter circuit is quite apparent but not objectionable. This amount of distortion will be apparent to the normal listener when 10 to 15 db of clipping is taking place. 2) Tune a selective communications receiver about 15 kc. to one side or the other of the frequency being transmitted. Use a short antenna or no antenna at all on the receiver so that the transmitter is not blocking the receiver. ( (3) Again with the normal talking into the microphone adjust the gain following the clipper -filter to the point where the side band splatter is being heard, and then slightly back off the gain after the clipper- filter until the splatter disappears. If the phase shift in the transmitter or modulator is not excessive the adjustment procedure given above will allow a clean signal. to be radiated regardless of any reasonable voice level being fed into the microphone. If a cathode -ray oscilloscope is available the modulated envelope of the transmitter should be checked with 30 to 70 cycle saw tooth waves on the horizontal axis. If the upper half of the envelope appears in general the same as the drawing of figure 15C, all is well and phase -shift is not excessive. However, if much more slope appears on the tops of the waves than is illustrated in this figure, it will be well to apply the second step in compensation in order to insure that sideband splatter cannot take place and to afford a still higher average percentage of modulation. This second step consists of the addition of a high -level splatter suppressor such as is illustrated in figure 17. www.americanradiohistory.com HANDBOOK T Splatter n MODULATOR 301 Suppression 5R4GY, 1616 836 Cz tit o z 111 C4 1/ 1 TO J i If 3000% WAVE FIL. TRANS INSULATED PLATE-MODULATED CLASS-C AMPLIFIER 7500 -10 000 OHMS LOAD FOR N.V. +B MOD. 115 +e R.F. FINAL V.A.C. Figure 17 HIGH -LEVEL SPLATTER SUPPRESSOR This circuit is effective in reducing splatter caused by negative -peak clipping in the modulated amplifier stage. The use of a two section filter as shown is recommended, although either a single m- derived or a con stant-k section may be used for greater economy. Suitable chokes, along with recommended capacitor values, are available from several manufacturers. 1000% WAVE 3001. Figure WAVE 16 ILLUSTRATING THE EFFECT OF PHASE SHIFT AND FILTERED WAVES OF DIFFERENT FREQUENCY Sketch (A) shows the effect of a clipper and a filter having a cutoff of about 3500 cycles on o wave of 3000 cycles. Note that no harmonics ore present in the wave so that phase shift following the clipper -filter will have no significant effect on the shape of the wave. (B) and (C) show the effect of phase shift on waves well below the cutoff frequency of the filter. Note that the "cant" placed upon the top of the wave causes the peak value to rise higher and higher above the clipping level as the frequency is lowered. It is for this reason that bass suppression before the clipper stage is desirable. Im- proved low -frequency response following the clipper -filter will reduce the phase shift and therefore the canting of the wave at the lower voice frequencies. The use of a high -level splatter suppressor after a clipper -filter system will afford the result shown in figure 18 since such a device will not permit the negative -peak clipping which the wave cant caused by audio -system phase shift can produce. The high -level splatter suppressor operates by virtue of the fact that it will not permit the plate voltage on the modulated amplifier to go completely to zero regardless of the incoming signal amplitude. Hence negative -peak clipping with its attendant splatter cannot take place. Such a device can, of course, also be used in a transmitter which does not incorporate a clipper- filter system. However, the full increase in average modulation level without serious distortion, afforded by the clipper- filter system, will not be obtained. A word of caution should be noted at this time in the case of tetrode final modulated amplifier stages which afford screen voltage modulation by virtue of a tap or a separate winding on the modulation transformer such as is shown in figure 9C of this chapter. If such a system of modulation is in use, the high -level splatter suppressor shown in figure 17 will not operate satisfactorily since negative -peak clipping in the stage can take place when the screen voltage goes too low. Clipper Circuits Two effective low -level clip- per- filter circuits are shown in figures 19 and 20. The circuit of figure 19 employs a 6J6 double triode as a clipper, each half of the 6J6 clipping one side of the impressed waveform. The optimum level at which the clipping operation begins is set by the value of the cathode resistor. A maximum of 12 to 14 db of clipping may be used with this circuit, which means that an extra 12 to 14 db of speech gain must precede the clipper. For a peak output of 8 volts from the clipper -filter, a peak audio signal of about 40 volts must be impressed upon the clipper input circuit. The 6C4 speech amplifier stage must therefore be considered as a part of the clipper circuit as www.americanradiohistory.com Amplitude Modulation 302 THE % 100 RADIO Figure 18 ACTION OF HIGH -LEVEL POS MODULATION SPLATTER SUPPRESSOR high -level splatter suppressor may be used in a transmitter without a clipper-filter to reduce negative -peak clipping, or such a unit may be used following a clipper filter to allow a higher average modulation level by eliminating the negative -peak clipping which the wave -cant caused by phase shift might produce. A ZERO AXIS 100 lb NEG. MODULATION SPLATTER- CAUSING NEGATIVE OVERMODULATION PEAK CUT OFF BY "NIGH -LEVEL SPLATTER SUPPRESSOR" the 12 to 14 db loss of gain incurred in the clipping process. A simple low pass filter made up of a 20 henry a.c. - d.c replacement type filter choke and two mica condensers follows the 616 clipper. This filter is designed for a cutoff frequency of about 3500 cycles when operating into a load impedance of % megohm. The output level of 8 volts peak is ample to drive a triode speech amplifier stage, such as a 6C4 or 6J5. A 6AL5 double diode series clipper is employed in the circuit of figure 20, and a commercially made low -pass filter is used to give somewhat better high frequency cutoff characteristics. A double triode is employed as a speech amplifier ahead of the clipper circuit. The actual performance of either circuit is about the same. To eliminate higher order products that may be generated in the stages following the clipper- filter, it is wise to follow the modulator with a high -level filter, as shown in figure 21. it compensates for Clipper Adjustment These clipper circuits have two adjustments: Adjust Gain and Adjust Clipping. The Adj. Gain control determines the modulation level of the transmitter. This control should be set so that over-modulation of the transmitter is impossible, regardless of the amount of clipping used. Once the Adj. Gain control has been roughly set, the Adj. Clip. control may be used to set the modulation level to any percentage below 100 %. As the modulation level is decreased, more and more clipping is introduced into the circuit, until a full 12 db of clipping is used. This means that the Adj. Gain control may be advanced some 12 db past the point where the clipping action started. Clipping action should start at 85% to 90% modulation when a sine wave is used for circuit adjustment purposes. Even though we may have cut off all frequencies above 3000 or 3500 cycles through the use of a filter system such as is shown in the circuits of figures 19 and 20, higher frequencies may again be introduced into the modulated wave by distortion in stages following the speech amplifier. Harmonics of the incoming audio frequencies may be generated in the driver stage for the modulator; they may be generated in the plate circuit of the modulator; or they may be generated by non -linearity in the modulated High -Level Filters amplifier itself. 6J6 6C4 6ÁU6 ADJUST CL/P. A7oULF ATAL MIC. SSK 20 N jSTANCOR Cilllj ADJUST GAIN AI 4.711 NEXT GRID TO 500 11 PEAK OUTPUT APPROX. /Z OE !V MAX W/TN OF CLIPPING. 5K, ALL RES /STOPS O.5 WATT UNLESS OTHERWISE MARKED ALL CAPACITORS /N {/E UNLESS OTHERWISE NOTED. K w Y250 v. Figure 19 CLIPPER FILTER USING 6J6 DOUBLE TRIODE STAGE www.americanradiohistory.com HANDBGOK Splatter 12AX7 6AL5 ADJUST GAIN Suppression 303 CHICAGO TRANS. LPF -2 FILTER TO NEXT GRID 4 7n SeK PEAK OUTPUTAPPROX 5V MAX. WITH 12 DB OF CLIPPING ADJUST SN,IW 00N CLIP. ALL RESISTORS 0.5 WATT UNLESS OTHERWISE MARKED ALL CAPACITORS IN OF UNLESS OTHERWISE MARKED. Figure 20 CLIPPER FILTER USING 6AL5 STAGE Regardless of the point in the system following the speech amplifier where the high audio frequencies may be generated, these frequencies can still cause a broad signal to be transmitted even though all frequencies above 3000 or 3500 cycles have been cut off in the speech amplifier. The effects of distortion in the audio system following the speech amplifier can be eliminated quite effectively through the use of a post- modulator filter. Such a filter must be used between the modulator plate circuit and the r -f amplifier which is being modulated. This filter may take three general forms in a normal case of a Class C amplifier plate modulated by a Class B modulator. The best method is to use a high level low -pass filter as CLASS C AMPLIFIER shown in figure 21 and discussed previously. Another method which will give excellent results in some cases and poor results in others, dependent upon the characteristics of the modulation transformer, is to "build out" the mod- ulation transformer into a filter section. This is accomplished as shown in figure 22 by placing mica capacitors of the correct value across the primary and secondary of the modulation transformer. The proper values for the capaci- CLASS C STAGE MODULATOR MODULATION TRANSFORMER B+ MOD. BI CLASS C Figure 22 "BUILDING -OUT" THE MODULATION Figure 21 ADDITIONAL HIGH -LEVEL LOW -PASS FILTER TO FOLLOW MODULATOR WHEN A LOW -LEVEL CLIPPER FILTER IS USED Suitable choke, along with recommended capacitor values, is available from several manufacturers. TRANSFORMER This expedient utilizes the leakage reactance of the modulation transformer in conjunction with the capacitors shown to make up a single- section low -pass filter. In order to determine exact values for CI and C2 plus C3, it is necessary to use a measurement setup such as Is shown in figure 23. However, experiment has shown in the case of a number of commercially available modulation transformers that a value for Cf of 0.002 -µfd. and C2 plus C3 of 0.004-µfd. will give satisfactory results. www.americanradiohistory.com 304 Amplitude Modulation , RADIO THE 1 11111 E%I/NÌ/111 AUDIO OSCILLATOR 1 I M 3íMIA11/1I1 4=11N11111 mri 1111111 oazA1/11111 2 -i. I111111 did Figure 23 TEST SETUP FOR BUILDING -OUT MODULATION TRANSFORMER Through the use of a test setup such as is shown and the method described in the text it is possible to determine the correct values for a specified filter characteristic in the built -out modulation transformer. tors C1 and C2 must, in the ideal case, be determined by trial and error. Experiment with a number of modulators has shown, however, that if a 0.002 pfd. capacitor is used for Cl and if the sum of C2 and C3 is made 0.004 tad.' (0.002 pfd. for CZ and 0.002 for C3) the ideal condition of cutoff above 3000 cycles will be approached in most cases with the "multiple match" type of modulation transformer. If it is desired to determine the optimum values of the capacitors across the transformer this can be determined in several ways, all of which require the use of a calibrated audio oscillator. One way is diagrammed in figure 23. The series resistors R1 and R2 should each be equal to V2 the value of the recommended plate to -plate load resistance for the Class B modulator tubes. Resistor R3 should be equal to the value of load resistance which the Class C modulated stage will present to the modulator. The meter V can be any type of a -c voltmeter. The indicating instrument on the secondary of the transformer can be either a cathode -ray oscilloscope or a high -impedance a -c voltmeter of the vacuum -tube or rectifier type. With a set -up as shown in figure 23 a plot of output voltage against frequency is made, at all times keeping the voltage across V constant, using various values of capacitance for C1 and C2 plus C3, When the proper values of capacitance have been determined which give substantially constant output up to about 3000 or 3500 cycles and decreasing output at all frequencies above, high -voltage mica capacitors can be substituted if receiving types were used in the tests and the transformer connected to the modulator and Class C amplifier. With the transformer reconnected in the transmitter a check of the modulated -wave output of the transmitter should be made using an audio oscillator as signal generator and an oscilloscope coupled to the transmitter output. With an input signal amplitude fed to the speech / WAWA Ap7/ON 11II min ,_/, -MI /i! 11 /I 100 200 300 11 RL - * 11111 11111 700 1000 500 2000 R,_. 3000 11 11 5000 FREQUENCY (CPS) Figure 24 BASE ATTENUATION CHART Frequency attenuation caused by various values of coupling capacitor with a grid resistor of 0.5 megohm in the following stage (Re > RL) amplifier of such amplitude that limiting does not take place, a substantially clean sine wave should be obtained on the carrier of the transmitter at all input frequencies up to the cutoff frequency of the filter system in the speech amplifier and of the filter which includes the modulation transformer. Above these cutoff frequencies very little modulation of the carrier wave should be obtained. To obtain a check on the effectiveness of the "built out" modulation transformer, the capacitors across the primary and secondary should be removed for the test. In most cases a marked deterioration in the waveform output of the modulator will be noticed with frequencies in the voice range from 500 to 1500 cycles being fed into the speech amplifier. A filter system similar to that shown in figure 17 may be used between the modulator and the modulated circuit in a grid -modulated or screen -modulated transmitter. Lower -voltage capacitors and low -current chokes may of course be employed. Boss Suppression Most of the power repre- sented by ordinary speech particularly the male voice) lies below 1000 cycles. If all frequencies below 400 or 500 cycles are eliminated or substantially attenuated, there is a considerable reduction in power but insignificant reduction in intelligi( www.americanradiohistory.com HANDBOOK bility. This means that the speech level may be increased considerably without overmodulation or overload of the audio system. In addition, if speech clipping is used, attenuation of the lower audio frequencies before the clipper will reduce phase shift and canting of the clipper output. A simple method of bass suppression is to reduce the size of the interstage coupling capacitors in a resistance coupled amplifier. Figure 24 shows the frequency characteristics caused by such a suppression circuit. A second simple bass suppression circuit is to place a small a.c. - d.c. type filter choke from grid to ground in a speech amplifier stage, as shown in figure 25. The systems described Distortion in the preceding paragraphs will have no effect in reducing a broad signal caused by non linearity in the modulated amplifier. Even though the modulating waveform impressed upon the modulated stage may be distortion free, if the modulated amplifier is non -linear distortion will be generated in the amplifier. The only way in which this type of distortion may be corrected is by making the modulated amplifier more linear. Degenerative feedback which includes the modulated amplifier in the loop will help in this regard. Plenty of grid excitation and high grid bias will go a long way toward making a plate modulated Class C amplifier linear, although such operating conditions will make more difficult the problem of TVI reduction. If this still does not give adequate linearity, the preceding buffer stage may be modulated 50 per cent or so at the same time and in the same phase as the final amplifier. The use of a grid leak to obtain the majority of the bias for a Class C stage will improve its linearity. The linearity of a grid -bias modulated r -f amplifier can be improved, after proper adjustments of excitation, grid bias, and antenna coupling have been made by modulating the stage which excites the grid -modulated amplifier. The preceding driver stage may be grid bias modulated or it may be plate modulated. Modulation of the driver stage should be in the same phase as that of the final modulated amplifier. Modulated Amplifier 15 -8 Suppression Bass The Bias -Shift Heising Modulator The simple Class A modulator is limited to an efficiency of about 30 %, and the tube must dissipate the full power input during periods of quiescence. Class AB and class B audio systems have largely taken the place of the old Heising modulator because of this great 305 .01 Figure 25 USE OF PARALLEL INDUCTANCE FOR BASS SUPPRESSION waste of power. It is possible, however, to vary the operating bias of the class A modulator in such a way as to allow class A operation only when an audio signal is applied to the grid of the tube. During resting periods, the bias can be shifted to a higher value, dropping the resting plate current and plate dissipation of the tube. When voice waveforms having o w average power are employed, the efficiency of the system is comparable to the popular class B modulator. 1 The characteristic curve for a class A modu- lator is shown in figure 26. Normal bias is used, and the operating point is placed in the middle of the linear portion of the Eg -Ip curve. Maximum plate input is limited by the plate dissipation of the tube under quiescent condition. The bias -shift modulator is biased close to plate current cut -off under no signal condition (figure 27). Resting plate current +IP , LINEAR PORTION -I oP Ec CURVE EG P All All! BIAS RESTING BIAS VOLTAGE P GRID INPUT SIGNAL Fìgur° 26 CHARACTERISTIC GRID VOLTAGE -PLATE CURRENT CURVE FOR CLASS A HEISING MODULATOR www.americanradiohistory.com SIGNAL +Ec CUT -OFF - PLATE OUTPUT /Inplitude Modulation 306 THE RADIO +te CLA S C AMPLIFIER DDI P AUDIO AMPLIFIER BIAS -SNIFT MODULATOR ^ (MAX 5/GNAL) Bf BIAS-SHIFT PLATE CURRENT EXCURSION REGT FIER I P (NO SIGNAL) +Ec EG FILTER --0 BIAS -SNIFT CONTROL TUBE CUT-OFF BIAS _ BIAS -SNIFT EXCURSION P NEGATIVE BIAS SUPPLY CLASS A OPERATING BIAS LINE QUIESCENT BIAS LINE Figure 27 BIAS -SHIFT MODULATOR Figure 28 BLOCK DIAGRAM OF BIAS -SHIFT MODULATOR OPERATING CHARACTERISTICS Modulator is biased close to plate current cut -off under no signal, B. Upon application of audio signal, the bias of the stage is shifted toward the class A operating point, A. Bias -shift voltage is obtained from audio condition, signal. and plate dissipation are therefore quite low. Upon application of an audio signal, the bias of the stage is shifted toward the class A operating point, preventing the negative peaks of the applied audio voltage from cutting off the plate current of the tube. As the audio voltage increases, the operating bias point is shifted to the right on figure 27 until the class A operating point is reached at maximum ex- citation. The bias -shift voltage may be obtained directly from the exciting signal by rectification, as shown in figure 28. A simple low pass filter system is used that will pass only the syllabic components of speech. Enough negative bias is applied to the bias -shift modulator to cut the resting plate current to the desired value, and the output of the bias control rectifier is polarized so as to "buck" the fixed bias voltage. No spurious modulation frequencies are generated, since the modulator operates class A throughout the audio cycle. This form of grid pulsing permits the modulator stage to work with an pverall efficiency of greater than 50 %, comparing favorably with the class B modulator. The expensive class B driver and output transformers are not required, since resistance coupling may be used in the input circuit of the bias -shift modulator, and a heavy -duty filter choke will serve as an impedance coupler for the modulated stage. Series and Parallel Control Circuits The bias -shift system make take one of several forms. A "series" control circuit is shown in figure 29. Resting bias is applied to the bias -shift modulator tube through the voltage divider R2 /R4. The bias control tube is placed across resistor R2. Quiescent bias for the modulator is set by adjusting R2. As the internal resistance of the bias control tube is varied at a syllabic rate the voltage drop across R2 will vary in unison. The modulator bias, therefore varies at the same rate. Excitation for the bias control tube is obtained from the audio signal through potentiometer RI which regulates the amplitude of the control signal. The audio signal is rectified by the bias control rectifier, and filtered by network R3 -Cl in the grid circuit of the bias con- trol tube. The "parallel" control system is illustrated in figure 30. Resting bias for the modulator is obtained from the voltage divider R2 /R4. Potentiometer R2 adjusts the resting bias level, determining the static plate current of the modulator. Resistor R3 serves as a bias resistor for the control tube, reducing its plate current to a low level. When an audio signal is applied via R1 to the grid of the control tube the internal resistance is lowered, decreasing the shunt resistance across R2. The negative modulator bias is therefore reduced. The bias axis of the modulator is shifted from the cut -off region to a point on the linear portion of the operating curve. The amount of bias -shift is controlled by the setting of potentiometer R1. Capacitor Cl in conjunction with bias resistor R3 form a syllabic filter for www.americanradiohistory.com HANDBOOK Heising Modulator BIAS-SHIFT SPEECH AMPLIFIER TO MODULATED B R-F AMPLIFIER MODULATOR 307 B4 SPEECH BIAS -SHIFT MODULATOR AMPLIFIER R TO MODULATED -F AMPLIFIER ADJUST OPERAT/N BIAS DJUST OPERA TING BIAS ADJUST RES T/NG BIAS BIAS CONTROL RECTIFIER BIAS CONTROL TUBE BIAS CONTROL TUBE NEGATIVE FtA MODULATOR BIAS NEGATIVE MODULATOR BIAS ADJUST REST /NG 81 S 2 Figure 29 "SERIES" CONTROL CIRCUIT Figure 30 FOR BIAS -SHIFT MODULATOR "PARALLEL" CONTROL The internal resistance of the bias control tube is varied of a syllabic rate to change the operating bias of the modulator CIRCUIT FOR BIAS -SHIFT MODULATOR The resistance to ground of point A in the bias network is varied at a syllabic rate by the bias control tube. tube. the control bias that is applied to the modu- lator stage. large value of plate dissipation is required for the bias -shift modulator tube. For plate voltages below 1500, the 211 (VT -4C) A may be used, while the 304 -TL is suitable for voltages up to 3000. As with normal class A amplifiers, low mu tubes function best in this circuit. www.americanradiohistory.com CHAPTER SIXTEEN Frequency Modulation and Radioteletype Transmission Exciter systems for FM and single sideband transmission are basically similar in that modification of the signal in accordance with the intelligence to be transmitted is normally accomplished at a relatively low level. Then the intelligence- bearing signal is amplified to the desired power level for ultimate transmission. True, amplifiers for the two types of signals are basically different; linear amplifiers of the Class, A or Class B type being used for ssb signals, while Class C or non -linear Class B amplifiers may be used for FM amplification. But the principle of low -level generation and subsequent amplification is standard for both types of transmission. 16 -1 the advantages of FM for certain types of communication pointed out. Since the distinguishing features of the two types of transmission lie entirely in the modulating circuits at the transmitter and in the detector and limiter circuits in the receiver, these parts of the communication system will receive the major portion of attention. Modulation is the process of altering a radio wave in accordance with the intelligence to be transmitted. The nature of the intelligence is of little importance as far as the process of modulation is concerned; it is the method by which this intelligence is made to give a distinguishing characteristic to the radio wave which will enable the receiver to convert it back into intelligence that determines the type of modulation being used. Figure 1 is a drawing of an r -f carrier amplitude modulated by a sine -wave audio voltage. After modulation the resultant modulated r-f wave is seen still to vary about the zero axis at a constant rate, but the strength of the individual r -f cycles is proportional to the amplitude of the modulation voltage. In figure 2, the carrier of figure 1 is shown frequency modulated by the same modulating voltage. Here it may be seen that modulation voltage of one polarity causes the carrier frequency to decrease, as shown by the fact that the individual r -f cycles of the carrier are spaced farther apart. A modulating voltage of the opposite polarity causes the frequency to Modulation Frequency Modulation The use of frequency modulation and the allied system of phase modulation has become of increasing importance in recent years. For amateur communication frequency and phase modulation offer important advantages in the reduction of broadcast and TV interference and in the elimination of the costly high -level modulation equipment most commonly employed with amplitude modulation. For broadcast work FM offers an improvement in signal -to-noise ratio for the high field intensities available in the local -coverage area of FM and TV broad- cast stations. In this chapter various points of difference between FM and amplitude modulation transmission and reception will be discussed and 308 www.americanradiohistory.com Frequency Modulation 309 UNMODULATED CARRIER AMPLITUDE f\, CARRIER SIDO FREQUENCY t SIDE FREQUENCY FREQUENCY Figure J FIGURE FIGURE 2 AM AND FM WAVES Figure 1 shows a sketch of the scope pattern of an amplitude modulated wave at the bottom. The center sketch shows the modulating wave and the upper sketch shows the carrier wave. Figure 2 shows at the bottom a sketch of a frequency modulated wave. In this case the center sketch also shows the modulating wave and the Laper sketch shows the carrier wave. Note that the carrier wave and the modulating wave are the same in either case, but that the waveform of the modulated wave is quite different in the two cases. increase, and this is shown by the r-f cycles being squeezed together to allow more of them to be completed in a given time interval. Figures 1 and 2 reveal two very important characteristics about amplitude- and frequency- modulated waves. First, it is seen that while the amplitude (power) of the signal is varied in AM transmission, no such variation takes place in FM. In many cases this advantage of FM is probably of equal or greater importance than the widely publicized noise reduction capabilities of the system. When 100 per cent amplitude modulation is obtained, the average power output of the transmitter must be increased by 50 per cent. This additional output must be supplied either by the modulator itself, in the high -level system, or by operating one or more of the transmitter stages at such a low output level that they are capable of producing the additional output without distortion, in the low -level system. On the other hand, a frequency -modulated transmitter requires an insignificant amount of power from the modulator and needs no provision for increased power output on modulation peaks. All of the stages between the oscillator and the antenna may be operated as high- efficiency Class B or Class C amplifiers or frequency multipliers. 3 AM SIDE FREQUENCIES For each AM modulating frequency, a pair of side frequencies is produced. The side frequencies are spaced away from the carrier by an amount equal to the modulation frequency, and their amplitude is directly proportional to the amplitude of the modulation. The amplitude of the carrier does not change under modulation. The second characteristic of FM and AM waves revealed by figures 1 and 2 is that both types of modulation result in distortion of the r -f carrier. That is, after modulation, the r -f cycles are no longer sine waves, as they would be if no frequencies other than the fundamental carrier frequency were present. It may be shown in the amplitude modulation case illustrated, that there are only two additional frequencies present, and these are the familiar side /requencies, one located on each side of the carrier, and each spaced from the carrier by a frequency interval equal to the modulation frequency. in regard to frequency and amplitude, the situation is as shown in figure 3. The strength of the carrier itself does not vary during modulation, but the strength of the side frequencies depends upon the percentage of modulation. At 100 per cent modulation the power in the side frequencies is equal to half that of the carrier. Under frequency modulation, the carrier wave again becomes distorted, as shown in figure 2. But, in this case, many more than two additional frequencies are formed. The first two of these frequencies are spaced from the carrier by the modulation frequency, and the additional side frequencies are located out on each side of the carrier and are also spaced from each other by an amount equal to the modulation frequency. Theoretically, there are an infinite number of side frequencies formed, but, fortunately, the strength of those beyond the frequency swing of the transmitter under modulation is relatively low. One set of side frequencies that might be formed by frequency modulation is shown in figure 4. Unlike amplitude modulation, the Carrier-Wove Distortion www.americanradiohistory.com 310 FM Transmission UNMODULATED CARRIER AMPLITUDE CARRIER SIDE FREQUENCIES IIli ill FREQUENCIES Ill FREQUENCY Figure 4 FM SIDE FREQUENCIES With FM each modulation frequency component causes a large number of side frequencies to be produced. The side frequencies are separated from each other and the carrier by an amount equal to the modulation frequency, but their anplitude varies greatly as the amount of modulation is changed. The carrier strength also varies greatly with frequency modulation. The side frequencies shown represent a case where the deviation each side of the "carrier" frequency is equal to five times the modulating frequency. Other amounts of deviation with the same modulation frequency would cause the relative strengths of the various sidebands to change widely. strength of the component at the carrier frequency varies widely in FM and it may even disappear entirely under certain conditions. The variation of strength of the carrier component is useful in measuring the amount of frequency modulation, and will be discussed in detail later in this chapter. One of the great advantages of FM over AM is the reduction in noise at the receiver which the system allows. If the receiver is made responsive only to changes in frequency, a considerable increase in signal -to -noise ratio is made possible through the use of FM, when the signal is of greater strength than the noise. The noise reducing capabilities of FM arise from the inability of noise to cause appreciable frequency modulation of the noise -plus -signal voltage which is applied to the detector in the receiver. Unlike amplitude modulation, the term percentage modulation means little in FM practice, unless the receiver characteristics are specified. There are, however, three terms, deviation, modulation index, and deviation ratio, which convey considerable information concerning the character of the FM wave. Deviation is the amount of frequency shift each side of the unmodulated carrier frequency which occurs when the transmitter is modulated. Deviation is ordinarily measured in kilocycles, and in a properly operating FM transFM Terms THE RADIO mitter it will be directly proportional to the amplitude of the modulating signal. When a symmetrical modulating signal is applied to the transmitter, equal deviation each side of the resting frequency is obtained during each cycle of the modulating signal, and the total frequency range covered by the FM transmitter is sometimes known as the suing. If, for instance, a transmitter operating on 1000 kc. has its frequency shifted from 1000 kc. to 1010 kc., back to 1000 kc., then to 990 kc., and again back to 1000 kc. during one cycle of the modulating wave, the deviation would be 10 kc. and the swing 20 kc. The modulation index of an FM signal is the ratio of the deviation to the audio modulating frequency, when both are expressed in the same units. Thus, in the example above if the signal is varied from 1000 kc. to 1010 kc. to 990 kc., and back to 1000 kc. at a rate (frequency) of 2000 times. a second, the modulation index would be 5, since the deviation (10 kc.) is 5 times the modulating frequency (2000 cycles, or 2 kc.). The relative strengths of the FM carrier and the various side frequencies depend directly upon the modulation index, these relative strengths varying widely as the modulation index is varied. In the preceding example, for instance, side frequencies occur on the high side of 1000 kc. at 1002, 1004, 1006, 1008, 1010, 1012, etc., and on the low frequency side at 998, 996, 994, 992, 990, 988, etc. In proportion to the unmodulated carrier strength (100 per cent), these side frequencies have the following strengths, as indicated by a modulation index of 5: 1002 and 998 -33 per cent, 1004 and 996-5 per cent, 1006 and 99436 per cent, 1008 and 992 -39 per cent, 1010 and 990 -26 per cent, 1012 and 988 -13 per cent. The carrier strength (1000 kc.) will be 18 per cent of its unmodulated value. Changing the amplitude of the modulating signal will change the deviation, and thus the modulation index will be changed, with the result that the side frequencies, while still located in the same places, will have different strength values from those given above. The deviation ratio is similar to the modulation index in that it involves the ratio between a modulating frequency and deviation. In this case, however, the deviation in question is the peak frequency shift obtained under full modulation, and the audio frequency to be considered is the maximum audio frequency to be transmitted. When the maximum audio frequency to be transmitted is 5000 cycles, for example, a deviation ratio of 3 would call for a peak deviation of 3 x 5000, or 15 kc. at full modulation. The noise -suppression capabilities of FM are directly related to the deviation ratio. As the deviation ratio is increased, www.americanradiohistory.com Narrow HANDBOOK the noise suppression becomes better if the signal is somewhat stronger than the noise. Where the noise approaches the signal in strength, however, low deviation ratios allow communication to be maintained in many cases where high- deviation -ratio FM and conventional AM are incapable of giving service. This assumes that a narrow -band FM receiver is in use. For each value of r -f signal -to -noise ratio at the receiver, there is a maximum deviation ratio which may be used, beyond which the output audio signal -to -noise ratio de- creases. Up to this critical deviation ratio, however, the noise suppression becomes progressively better as the deviation ratio is in- creased. For high- fidelity FM broadcasting purposes, deviation ratio of 5 is ordinarily used, the maximum audio frequency being 15,000 cycles, and the peak deviation at full modulation being 75 kc. Since a swing of 150 kc. is covered by the transmitter, it is obvious that wide band FM transmission must necessarily be confined to the v -h -f range or higher, where room for the signals is available. In the case of television sound, the deviation ratio is 1.67; the maximum modulation frequency is 15,000 cycles, and the transmitter deviation for full modulation is 25 kc. The sound carrier frequency in a standard TV signal is located exactly 4.5 Mc. higher than the picture carrier frequency. In the intercarrier TV sound system, which recently has become quite widely used, this constant difference between the picture carrier and the sound carrier is employed within the receiver to obtain an FM sub-carrier at 4.5 Mc. This 4.5 Mc. sub-carrier then is demodulated by the FM detector to obtain the sound signal which accompanies the picture. a Narrow -band Narrow -Band FM t r an s- mission has become standardized for use by the mobile services such as police, fire, and taxicab communication, and also on the basis of a temporary authorization for amateur work in portions of each of the amateur radiotelephone bands. A maximum deviation of 15 kc. has been standardized for the mobile and commercial communication services, while a maximum deviation of 3 kc. is authorized for amateur NBFM communication. the transmitter is most of them may mission, when a (speech or music) Band FM 311 swung are so small that be ignored. In FM transcomplex modulating wave is used, still additional side frequencies resulting from a beating together of the various frequency components in the modulating wave are formed. This is a situation that does not occur in amplitude modulation and it might be thought that the large number of side frequencies thus formed might make the frequency spectrum produced by an FM transmitter prohibitively wide. Analysis shows, however, that the additional side frequencies are of very small amplitude, and, instead of increasing the bandwidth, modulation by a complex wave actually reduces the effective bandwidth of the FM wave. This is especially true when speech modulation is used, since most of the power in voiced sounds is concentrated at low frequencies in the vicinity of 400 cycles. The bandwidth required in an FM receiver is a function of a number of factors, both theoretical and practical. Basically, the bandwidth required is a function of the deviation ratio and the maximum frequency of modulation, although the practical consideration of drift and ease of receiver tuning also must be considered. Shown in figure 5 are the frequency spectra (carrier and sideband frequencies) associated with the standard FM broadcast signal, the TV sound signal, and an amateur band narrow -band FM signal with full modulation using the highest permissible modulating frequency in each case. It will be seen that for low deviation ratios the receiver bandwidth should be at least four times the maximum frequency deviation, but for a deviation ratio of 5 the receiver bandwidth need be only about 2.5 times the maximum frequency deviation. FM Transmission Bandwidth Required by FM As the above discussion has indicated, many side frequencies are set up when a radio - frequency carrier is frequency modulated; theoretically, in fact, an infinite number of side frequencies is formed. Fortunately, however, the amplitudes of those side frequencies falling outside the frequency range over which 16 -2 Direct FM Circuits Frequency modulation may be obtained either direct method, in which the frequency of an oscillator is changed directly by the modulating signal, or by the indirect method which makes use of phase modulation. Phase modulation circuits will be discussed in section 16 -3. A successful frequency modulated transmitter must meet two requirements: (1) The frequency deviation must be symmetrical about a fixed frequency, for symmetrical modulation voltage. (2) The deviation must be directly proportional to the amplitude of the modulation, and independent of the modulation frequency. There are several methods of direct frequency modulation which will fulfill these by the www.americanradiohistory.com 312 Transmission FM OA FM BROADCAST THE AUDIO IN DEVIATION - 75 KC. MOD. FREQ.-15 KC. MOD. INDEX - 5 e Ñ Iy l IS .11 -105 -90 -75 -60 -45 -30 -15 V 1! I! C2 47 R,10K y } R n OSCILLATOR IN 1.75 MC. RANGE 6BA6 1 RADIO 100K f ^ +15 +30 +45 +60 +75 +90 +105 -T- 00 K- C31., 1 POT. 470 K 1 O TV SOUND ati R ,r Ei; -4514C © KC. MOO. FREQ.- 15 KC. MOD. INDEX -1.67 -30KC. -15 Vlt fl +15 KC IN KC 4-30 KC +45 KC. DEVIATION - 3 KC. MOD. EREQ.- 3 KC. AMATEUR NBFM MOD. INDEX < -6KC. -3KC. + - cillator tank is varied. By applying audio I I 3 KC. CENTER FREQUENCY Figure 5 EFFECT OF FM MODULATION INDEX Showing the side - frequency amplitude and distribution for the three most conrnon modulation indices used in FM work. The maximum modulating frequency and maximum deviation are shown in each case. requirements. Some of these methods described in the following paragraphs. Reactance-Tube Modulators Figure 6 REACTANCE -TUBE MODULATOR This circuit is convenient for direct frequency modulation of on oscillator in the 1.75 -Mc. range. Capacitor C, may be only the input capacitance of the tube, or a small trimmer capacitor may be included to permit a variation in the sensitivity of the reactance tube. II* fl .0068 +150 -200V. REGULATED DEVIATION- 25 will be practical One of the most ways of obtaining direct fre- quency modulation is through the use of a reactance -tube modulator. In this arrangement the modulator plate- cathode circuit is connected across the oscillator tank circuit, and made to appear as either a capacitive or inductive reactance by exciting the modulator grid with a voltage which either leads or lags the oscillator tank voltage by 90 degrees. The leading or lagging grid voltage causes a corresponding leading or lagging plate current, and the plate- cathode circuit appears as a capacitive or inductive reactance across the oscillator tank circuit. When the transconductance of the modulator tube is varied, by varying one of the element voltages, the magnitude of the reactance across the os- mod- ulating voltage to one of the elements, the transconductance, and hence the frequency, may be varied at an audio rate. When properly designed and operated, the reactance -tube modulator gives linear frequency modulation, and is capable of producing large amounts of deviation. There are numerous possible configurations of the reactance -tube modulator circuit. The difference in the various arrangements lies principally in the type of phase -shifting circuit used to give a grid voltage which is in phase quadrature with the r -f voltage at the modulator plate. Figure 6 is a diagram of one of the most popular forms of reactance -tube modulators. The modulator tube, which is usually a pentode such as a 6BA6, 6ÁU6, or 6CL6, has its plate coupled through a blocking capacitor, to the "hot" side of the oscillator grid circuit. Another blocking capacitor, C2, feeds r.f. to the phase shifting network R -C, in the modulator grid circuit. If the resistance of R is made large in comparison with the reactance of C, at the oscillator frequency, the cur- C through the R -C, combination will be nearly in phase with the voltage across the tank circuit, and the voltage across C, will lag the oscillator tank voltage by almost 90 degrees. The result of the 90- degree lagging voltage on the modulator grid is that its plate current lags the tank voltage by 90 degrees, and the reactance tube appears as an inductance in shunt with the oscillator inductance, thus raising the oscillator frequency. The phase- shifting capacitor C, can consist of the input capacitance of the modulator tube and stray capacitance between grid and ground. rent www.americanradiohistory.com Reactance Tube HANDBOOK 313 in figures 6 and 7. The audio input may be applied to the suppressor grid, rather than the AUDIO IN +150 -2D0 V. REGULATED Figure 7 ALTERNATIVE REACTANCE -TUBE MODULATOR This circuit is often preferable for use in the lower frequency range, although it may be used at 1.75 Mc. and above if desired. In the schematic above the reactance tube is shown connected across the voltage- divider capacitors of a Clapp oscillator, although the modulator circuit may be used with any common type of oscillator. However, better control of the operating conditions of the modulator may be had through the use of a variable capacitor as C,. Resistance R will usually have a value of between 4700 and 100,000 ohms. Either resistance or transformer coupling may be used to feed audio voltage to the modulator grid. When a resistance coupling is used, it is necessary to shield the grid circuit adequately, since the high impedance grid circuit is prone to pick up stray r -f and low frequency a -c voltage, and cause undesired frequency modulation. An alternative reactance modulator circuit is shown in figure 7. The operating conditions are generally the same, except that the r -f excitation voltage to the grid of the reactance tube is obtained effectively through reversing the R and C, of figure 6. In this circuit a small capacitance is used to couple r.f. into the grid of the reactance tube, with a relatively small value of resistance from grid to ground. This circuit has the advantage that the grid of the tube is at relatively low impedance with respect to r.f. However, the circuit normally is not suitable for operation above a few megacycles due to the shunting capacitance within the tube from grid to ground. Either of the reactance -tube circuits may be used with any of the common types of oscillators. The reactance modulator of figure 6 is shown connected to the high- impedance point of a conventional hot -cathode Hartley oscillator, while that of figure 7 is shown connected across the low- impedance capacitors of a series -tuned Clapp oscillator. There are several possible variations of the basic reactance-tube modulator circuits shown control grid, if desired. Another modification is to apply the audio to a grid other than the control grid in a mixer or pentagrid converter tube which is used as the modulator. Generally, it will be found that the transconductance variation per volt of control -element voltage variation will be greatest when the control (audio) voltage is applied to the control grid. In cases where it is desirable to separate completely the audio and r -f circuits, however, applying audio voltage to one of the other elements will often be found advantageous despite the somewhat lower sensitivity. One of the simplest methods of adjusting the phase shift to the correct amount is to place a pair of earphones in series with the oscillator cathode -to- ground circuit and adjust the phase -shift network until minimum sound is heard in the phones when frequency modulation is taking place. If an electron -coupled or Hartley oscillator is used, this method requires that the cathode circuit of the oscillator be inductively or capacitively coupled to the grid circuit, rather than tapped on the grid coil. The phones should be adequately bypassed for r.f. of course. Adjusting the Phase Shift Stabilization Due to the presence of the react- ance-tube frequency modulator, the stabilization of an FM oscillator in regard to voltage changes is considerably more involved than in the case of a simple self-controlled oscillator for transmitter frequency control. If desired, the oscillator itself may be made perfectly stable under voltage changes, but the presence of the frequency modulator destroys the beneficial effect of any such stabilization. It thus becomes desirable to apply the stabilizing arrangement to the modulator as well as the oscillator. If the oscillator itself is stable under voltage changes, it is only necessary to apply voltage- frequency compensation to the modulator. Reactance -Tube Modulators Two simple reactance -t u b e modulators that may be applied to an existing v.f.o. are illustrated in figures 8 and 9. The circuit of figure 8 is extremely simple, yet effective. Only two tubes are used exclusive of the voltage regulator tubes which perhaps may be already incorporated in the v.f.o. A 6AU6 serves as a high -gain voltage amplifier stage, and a 6CL6 is used as the reactance modulator since its high value of transconductance will permit a large value of lagging current to be drawn under modulation swing. The unit should be www.americanradiohistory.com FM 314 Transmission RADIO THE 6AU6 (r 6CL6 50 L U 68 ULF 4.7 GRFi D OR RFC CATHODE NOTE: 2.814H ALL RESISTORS 0.5 WATT UNLESS OTHERWISE NOTED ALL CAPACITORS IN 1./F UNLESS OTHERWISE NOTED VR ADJUST FOR CORRECT CURRENT Figure 8 SIMPLE FM REACTANCE -TUBE MODULATOR mounted in close proximity to the v.f.o. so that the lead from the 6CL6 to the grid circuit of the oscillator can be as short as possible. A practical solution is to mount the reactance modulator in a small box on the side of the v -f -o cabinet. By incorporating speech clipping in the reactance modulator unit, a much more effective use is made of a given amount of deviation. When the FM signal is received on an AM receiver by means of slope detection, the use of speech clipping will be noticed by the greatly increased modulation level of the FM signal, and the attenuation of the center frequency null of no modulation. In many cases, it is difficult to tell a speech -clipped FM signal from the usual AM signal. A more complex FM reactance modulator incorporating a speech clipper is shown in figure 9. A 12AX7 double triode speech amplifier provides enough gain for proper clipper action when a high level crystal microphone is used. A double diode 6AL5 speech clipper is used, the clipping level being set by the potentiometer controlling the plate voltage applied to the diode. A 6CL6 serves as the reactance modu- lator. 12AX7 The reactance modulator may best be adjusted by listening to the signal of the v -f -o exciter at the operating frequency and adjusting the gain and clipping controls for the best modulation level consistent with minimum side band splatter. Minimum clipping occurs when the Adj. Clip. potentiometer is set for mayimum voltage on the plates of the 6AL5 clipper tube. As with the case of all reactance modulators, a voltage regulated plate supply is required. Linearity Test It is almost a necessity to run a static test on the reactance tube frequency modulator to determine its linearity and effectiveness, since small changes in the values of components, and in stray capacitances will almost certainly alter the modulator characteristics. A frequency- versus -control -voltage curve should be plotted to ascertain that equal increments in control voltage, both in a positive and a negative direction, cause equal changes in frequency. If the curve shows that the modulator has an appreciable amount of non -linearity, changes in bias, electrode voltages, r -f excitation, and resistance 6AL5 DJUST GAIN 6CL6 500 ULF ri 66LÚ CHICAGO TRANS. 4.7A LPF -2 FILTER TO GRID OR CAT/IODE OF vFO. R FC 2.SMH 5 .01 .01 NOTE'. ALL CAPACITORS IN OF UNLESS OTHERWISE NOTED ALL RESISTORS 0.5 WATT UNLESS OTHERWISE NOTED ADJUST 100A CLIPPING Figure 9 FM REACTANCE MODULATOR WITH SPEECH CLIPPER www.americanradiohistory.com f ADJUST FOR CORRECT VR CURRENT e+ Phase HANDBOOK TO MODULATOR CONTROL ELEMENT values may be made to obtain a straight -line characteristic. Figure 10 shows a method of connecting two 4/2 -volt C batteries and a potentiometer to plot the characteristic of the modulator. It will be necessary to use a zero-center voltmeter to measure the grid voltage, or else reverse the voltmeter leads when changing from positive to negative grid voltage. When a straight -line characteristic for the modulator is obtained by the static test method, the capacitances of the various by -pass capacitors in the circuit must be kept small to retain this characteristic when an audio voltage is used to vary the frequency in place of the d -c voltage with which the characteristic was plotted. 16 -3 Phase Modulation By means of phase modulation (PM) it is possible to dispense with self -controlled oscillators and to obtain directly crystal -controlled FM. In the final analysis, PM is sim- ply frequency modulation in which the deviation is directly proportional to the modulation frequency. If an audio signal of 1000 cycles causes a deviation of % kc., for example, a 2000 -cycle modulating signal of the same amplitude will give a deviation of 1 kc., and so on. To produce an FM signal, it is necessary to make the deviation independent of the modulation frequency, and proportional only to the modulating signal. With PM this is done by including a frequency correcting network in the transmitter. The audio correction network must have an attenuation that varies directly with frequency, and this requirement is easily met by a very simple resistance- capacity network. The only disadvantage of PM, as compared to direct FM such as is obtained through the use of a reactance -tube modulator, is the fact that very little frequency deviation is produced directly by the phase modulator. The deviation produced by a phase modulator is independent of the actual carrier frequency on 315 which the modulator operates, but is dependent only upon the phase deviation which is being produced and upon the modulation frequency. Expressed as an equation: modulating frequency Where Fd is the frequency deviation one way from the mean value of the carrier, and M, is the phase deviation accompanying modulation expressed in radians(a radian is approximately 57.3 °). Thus, to take an example, if the phase deviation is % radian and the modulating frequency is 1000 cycles, the frequency deviation applied to the carrier being passed through the phase modulator will be 500 cycles. It is easy to see that an enormous amount of multiplication of the carrier frequency is required in order to obtain from a phase modulator the frequency deviation of 75 kc. required for commercial FM broadcasting. However, for amateur and commercial narrow -band FM work (NBFM) only a quite reasonable number of multiplier stages are required to obtain a deviation ratio of approximately one. Actually, phase modulation of approximately one -half radian on the output of a crystal oscillator in the 80 -meter band will give adequate deviation for 29 -Mc. NBFM radiotelephony. For example; if the crystal frequency is 3700 kc., the deviation in phase produced is t/ radian, and the modulating frequency is 500 cycles, the deviation in the 80 -meter band will be 250 cycles. But when the crystal frequency is multiplied on up to 29,600 kc. the frequency deviation will also be multiplied by 8 so that the resulting deviation on the 10 -meter band will be 2 kc. either side of the carrier for a total swing in carrier frequency of 4 kc. This amount of deviation is quite adequate for NBFM work. Odd -harmonic distortion is produced when FM is obtained by the phase- modulation method, and the amount of this distortion that can be tolerated is the limiting factor in determining the amount of PM that can be used. Since the aforementioned frequency- correcting network causes the lowest modulating frequency to have the greatest amplitude, maximum phase modulation takes place at the lowest modulating frequency, and the amount of distortion that can be tolerated at this frequency determines the maximum deviation that can be obtained by the PM method. For high -fidelity broadcasting, the deviation produced by PM is limited to an amount equal to about one -third of the lowest modulating frequency. But for NBFM work the deviation may be as high as 0.6 of the modulating frequency before distortion becomes objectionable on voice modulation. In other terms this means that phase deviations as high as 0.6 radian may be used for amateur and commercial NBFM transmission. Fd = Figure 10 REACTANCE -TUBE LINEARITY CHECKER Modulation www.americanradiohistory.com MP 316 FM REACTANCE TUBE Transmission CRYSTAL OSCILLATOR TUBE r -- THE y RADIO When a single- frequency mod ulating voltage is used with an FM transmitter, the relative amplitudes of the various sidebands and the carrier vary widely as the deviation is varied by increasing or decreasing the amount of modulation. Since the relationship between the amplitudes of the various sidebands and carrier to the audio modulating frequency and the deviation is known, a simple method of measuring the deviation of a frequency modulated transmitter is possible. In making the measurement, the result is given in the form of the modulation index for a certain amount of audio input. As previously described, the modulation index is the ratio of the peak frequency deviation to the frequency of the audio modulation. The measurement is made by applying a sine -wave audio voltage of known frequency to the transmitter, and increasing the modulation until the amplitude of the carrier component of the frequency modulated wave reaches zero. The modulation index for zero carrier may then be determined from the table below. As may be seen from the table, the first point of zero carrier is obtained when the modulation Measurement NE %T STAGE LOW-C AUDIO Figure 11 REACTANCE -TUBE MODULATION OF CRYSTAL OSCILLATOR STAGE A simple reactance modula cor normally used for FM may also be used for PM by connecting it to the plate circuit of a crystal oscillator stage as shown in figure 11. Phase -Modulation Circuits Another PM circuit, suitable for operation on 20, 15 and 10 meters with the use of 80 meter crystals is shown in figure 12. A double triode 12AX7 is used as a combination Pierce crystal oscillator and phase modulator. C, should not be thought of as a neutralizing condenser, but rather as an adjustment for the phase of the r -f voltage acting between the grid and plate of the 12AX7 phase modulator. C2 acts as a phase angle and magnitude control, and both these condensers should be adjusted for maximum phase modulation capabili- ties of the circuit. Resonance of the circuit is established by the iron slug of coil L, -L,. A 6CL6 is used as a doubler to 7 Mc. and delivers approximately 2 watts on this band. Additional doubler stages may be added after the 6CL6 stage to reach the desired band of operation. Still another PM circuit, which is quite widely used commercially, is shown in figure 13. In this circuit L and C are made resonant at a frequency which is 0.707 times the operating frequency. Hence at the operating frequency the inductive reactance is twice the capacitive reactance. A cathode follower tube acts as a variable resistance in series with the L and C which go to make up the tank circuit. The operating point of the cathode follower should be chosen so that the effective resistance in series with the tank circuit (made up of the resistance of the cathode- follower tube in parallel with the cathode bias resistor of the cathode follower) is equal to the capacitive reactance of the tank capacitor at the operating frequency. The circuit is capable of about plus or minus % radian deviation with tolerable distortion. of Deviation index has a value of 2.405, -in other words, when the deviation is 2.405 times the modulation frequency. For example, if a modulation frequency of 1000 cycles is used, and the modulation is increased until the first carrier null is obtained, the deviation will then be 2.405 times the modulation frequency, or 2.405 kc. If the modulating frequency happened to be 2000 cycles, the deviation at the first null would be 4.810 kc. Other carrier nulls will be obtained when the index is 5.52, 8.654, and at increasing values separated approximately by rr. The following is a listing of the modulation index at successive carrier nulls up to the tenth: Zero carrier Modulation point no. index 1 2 3 4 5 6 7 8 9 10 2.405 5.520 8.654 11.792 14.931 18.071 21.212 24353 27.494 30.635 The only equipment required for making the measurements is a calibrated audio oscillator of good wave form, and a communication receiver equipped with a beat oscillator and crystal filter. The receiver should be used with its crystal filter set for minimum bandwidth to exclude sidebands spaced from the carrier by the modulation frequency. The un- www.americanradiohistory.com HANDBOOK FM Reception 317 6C L6 12AX7 TO DOUBLER STAGES 250 100F L -SET RSl E. }SPACED 71 APART ON 36E. r POWDERED IRON COR L 2-Is T. I L3 -377'. H20E. CLOSE-SPACED I NOTE. ALL RESISTORS 0.5 WA rr FORM O/A. B+ 300 V. UNLESS OTHERWISE NOTED ALL CAPACITORS IN 4/F UNLESS OTHERWISE NOTED Figure 12 REACTANCE MODULATOR FOR 10, 15 AND modulated carrier is accurately tuned in on the receiver with the beat oscillator operating. Then modulation from the audio oscillator is applied to the transmitter, and the modulation is increased until the first carrier null is obtained. This carrier null will correspond to a modulation index of 2.405, as previously mentioned. Successive null points will correspond to the indices listed in the table. A volume indicator in the transmitter audio system may be used to measure the audio level required for different amounts of deviation, and the indicator thus calibrated in terms of frequency deviation. If the measurements are made at the fundamental frequency of the oscillator, it will be necessary to multiply the frequency deviation by the harmonic upon which the transmitter is operating, of course. It will probably be most convenient to make the determination at some frequency intermediate between that of the oscillator and that at which the transmitter is operating, and then to multiply the result by the frequency multiplication between that frequency and the transmitter output frequency. 16 -4 A Reception of FM Signals conventional communications receiver may be used to receive narrow -band FM transmissions, although performance will be much poorer than can be obtained with an NBFM receiver or adapter. However, a receiver specifically designed for FM reception must be used when it is desired to receive high deviation FM such 20 METER OPERATION as used by FM broadcast stations, TV sound, and mobile communications FM. The FM receiver must have, first of all, a bandwidth sufficient to pass the range of frequencies generated by the FM transmitter. And since the receiver must be a superheterodyne if it is to have good sensitivity at the frequencies to which FM is restricted, i-f bandwidth is an important factor in its design. The second requirement of the FM receiver is that it incorporate some sort of device for converting frequency changes into amplitude changes, in other words, a detector operating on frequency variations rather than amplitude variations. The third requirement, and one which is necessary if the full noise reducing capa- 65.17 R F. PHASE- MODULATED OUTPUT INPUT fo +B 200 V. 01 AUDIO IN XL ABOUT XC ABOUT 1500n AT T50 R. AT fo f0 Figure 13 CATHODE -FOLLOWER PHASE MODULATOR The phase modulator illustrated above is quite satisfactory when the stage is to be operated on a single frequency or over a narrow range of frequencies. www.americanradiohistory.com 318 MIXE T FM I. r. AMPLIFIER Transmission LIMITER FREQUENCY DETECTOR (DISCRIMINATOR) THE RADIO AUDIO AMP. OSCILLATOR F Figure 14 FM RECEIVER BLOCK DIAGRAM Up to the amplitude limiter stage, the FM receiver is similar to an AM receiver, except for a somewhat wider i -f bandwidth. The limiter removes any amplitude modulation, and the frequency detector following the limiter converts frequency variations into amplitude variations. bilities of the FM system of transmission are a limiting device to eliminate amplitude variations before they reach the detector. A block diagram of the essential parts of an FM receiver is shown in figure 14. desired, is The simplest device for con venting frequency variations to amplitude variations is an "off- tune" resonant circuit, as illustrated in figure 15. With the carrier tuned in at point "A," a certain amount of r -f voltage will be developed across the tuned circuit, and, as the frequency is varied either side of this frequency by the modulation, the r -f voltage will increase and decrease to points "C" and "B" in accordance with the modulation. If the voltage across the tuned circuit is applied to an The Frequency Detector ordinary detector, the detector output will vary in accordance with the modulation, the amplitude of the variation being proportional to the deviation of the signal, and the rate being equal to the modulation frequency. It is obvious from figure 15 that only a small portion of the resonance curve is usable for linear conversion R E Q U E N C Y Figure 15 SLOPE DETECTION OF FM SIGNAL One side of the response characteristic of a tuned circuit or of on i -f amplifier may be used as shown to convert frequency variations of an incoming signal into amplitude variations. of frequency variations into amplitude variations, since the linear portion of the curve is rather short. Any frequency variation which exceeds the linear portion will cause distortion of the recovered audio. It is also obvious by inspection of figure 15 that an AM receiver used in this manner is wide open to signals on the peak of the resonance curve and also to signals on the other side of the resonance curve. Further, no noise limiting action is afforded by this type of reception. This system, therefore, is not recommended for FM reception, although widely used by amateurs for occasional NBFA1 reception. Travis Discriminator Another form of frequency detector or discriminator, is shown in figure 16. In this arrangement two tuned circuits are used, one tuned on each side of the i -f amplifier frequency, and with their resonant frequencies spaced slightly more than the expected transmitter swing. Their outputs are combined in a differential rectifier so that the voltage across the series load resistors, R, and R2, is equal to the algebraic sum of the individual output voltages of each rectifier. When a signal at the At its "center" frequency t he discriminator n produces zero output voltage. either side of Figure 16 TRAVIS DISCRIMINATOR This type of discriminator makes use two off-tuned resonant circuits coupled to of a single primary winding. The circuit is capable of excellent linearity, but is difficult to align. On this frequency It gives a voltage of a polarity and magnitude which depend on the direction and amount FREQUENCY of frequency shift. Figure 17 DISCRIMINATOR VOLTAGE -FREQUENCY CURVE www.americanradiohistory.com FM HANDBOOK Figure 18 FOSTER-SEELEY DISCRIMINATOR This discriminator is the most widely used circuit since it is capable of excellent linearity and is relatively simple to align when proper test equipment is available. i -f mid -frequency is received, the voltages across the load resistors are equal and opposite, and the sum voltage is zero. As the r -f signal varies from the mid -frequency, however, these individual voltages become unequal, and a voltage having the polarity of the larger voltage and equal to the difference between the two voltages appears across the series resistors, and is applied to the audio amplifier. The relationship between frequency and discriminator output voltage is shown in figure 17. The separation of the discriminator peaks and the linearity of the output voltage vs. frequency curve depend upon the discriminator frequency, the Q of the tuned circuits, and the value of the diode load resistors. As the intermediate (and discriminator) frequency is increased, the peaks must be separated further to secure good linearity and output. Within limits, as the diode load resistance or the Q is reduced, the linearity improves, and the separation between the peaks must be greater. The most widely used form of discriminator is that shown in figure 18. This type of discriminator yields an output- voltage- versus -frequency characteristic similar to that shown in figure 19. Here, again, the output voltage is equal to the algebraic sum of the voltages developed across the load resistors of the two diodes, the resistors being connected in series to ground. However, this Foster-Seeley discriminator requires only two tuned circuits instead of the three used in the previous discriminator. The operation of the circuit results from the phase relationships existing in a transformer having a tuned secondary. In effect, as a close examination of the circuit will reveal, the primary circuit is in series, for r.f., with each half of the secondary to ground. When the received signal is at the resonant frequency of the secondary, the r -f voltage across the secondary is 90 degrees out of phase with that across the primary. Since each diode is connected across one half of the secondary windFoster -Seeley Discriminator Reception 319 SECONDARY VOLTAGE Figure 19 DISCRIMINATOR VECTOR DIAGRAM A signal at the resonant frequency of the secondary will cause the secondary voltage to be 90 degrees out of phase with the primary voltage, as shown at A, and the resultant voltages R and R' are equal. If the signal frequency changes, the phase relationship also changes, and the resultant voltages are no longer equal, as shown at B. A differ ential rectifier is used to give an output voltage proportional to the difference between R and R'. ing and the primary winding in series, the resultant r -f voltages applied to each are equal, and the voltages developed across each diode load resistor are equal and of opposite polarity. Hence, the net voltage between the top of the load resistors and ground is zero. This is shown vectorially in figure 19A where the resultant voltages R and R which are applied to the two diodes are shown to be equal when the phase angle between primary and secondary voltages is 90 degrees. If, however, the signal varies from the resonant frequency, the 90- degree phase relationship no longer exists between primary and secondary. The result of this effect is shown in figure 1913 where the secondary r -f voltage is no longer 90 degrees out of phase with respect to the primary voltage. The resultant voltages applied to the two diodes are now no longer equal, and a d -c voltage proportional to the difference between the r -f voltages applied to the two diodes will exist across the series load resistors. As the signal frequency varies back and forth across the resonant frequency of the discriminator, an a -c voltage of the same frequency as the original modulation, and proportional to the deviation, is developed and passed on to the audio amplifier. Ratio Detector One of the more recent types of FM detector circuits, called the ratio detector is diagrammed in figure 20. The input transformer can be designed so that the parallel input voltage to the diodes can be taken from a tap on the primary of the trans- 320 FM .0001 Transmission THE RADIO RFC 6SJ 7 TO 015C R I M- INATOR +250 A.F. OUTPUT Figure 20 RATIO DETECTOR CIRCUIT The parallel voltage to the diodes in a ratio detector may be obtained from a tap on the primary winding of the transformer or from o third winding. Note that one of the diodes is reversed from the system used with the Foster -Seeley discriminator, and that the output circuit is completely different. The ratio detector does not have to be preceded by o limiter, but is more difficult to align for distortion -free output than the conventional discriminator. former, or this voltage may be obtained from a tertiary winding coupled to the primary. The r -f choke used must have high impedance at the intermediate frequency used in the receiver, although this choke is not needed if the transformer has a tertiary winding. The circuit of the ratio detector appears very similar to that of the more conventional discriminator arrangement. However, it will be noted that the two diodes in the ratio detector are poled so that their d -c output voltages add, as contrasted to the Foster -Seeley circuit wherein the diodes are poled so that the d-c output voltages buck each other. At the center frequency to which the discriminator transformer is tuned the voltage appearing at the top of the 1- megohm potentiometer will be one half the d -c voltage appearing at the a -v-c output terminal -since the contribution of each diode will be the same. However, as the input frequency varies to one side or the other of the tuned value (while remaining within the pass band of the i -f amplifier feeding the detector) the relative contributions of the two diodes will be different. The voltage appearing at the top of the 1- megohm volume control will increase for frequency deviations in one direction and will decrease for frequency deviations in the other direction from the mean or tuned value of the transformer. The audio output voltage is equal to the ratio of the relative contributions of the two diodes, hence the name ratio detector. The ratio detector offers several advantages over the simple discriminator circuit. The circuit does not require the use of a limiter preceding the detector since the circuit is inherently insensitive to amplitude modulation on Figure 21 LIMITER CIRCUIT One, or sometimes two, limiter stages normally precede the discriminator so that a constant signal level will be fed to the FM detector. This procedure eliminates amplitude variations in the signal fed to the discriminator, so that it will respond only to frequency changes. an incoming signal. This factor alone means that the r -f and i -f gain ahead of the detector can be much less than the conventional discriminator for the same overall sensitivity. Further, the circuit provides a -v -c voltage for controlling the gain of the preceding r -f and i -f stages. The ratio detector is, however, susceptible to variations in the amplitude of the incoming signal as is any other detector circuit except the discriminator with a limiter preceding it, so that a -v -c should be used on the stages preceding the detector. Limiters The limiter of an FM receiver using a conventional discriminator serves to remove amplitude modulation and pass on to the discriminator a frequency modulated signal of constant amplitude; a typical circuit is shown in figure 21. The limiter tube is operated as an i -f stage with very low plate voltage and with grid leak bias, so that it overloads quite easily. Up to a certain point the output of the limiter will increase with an increase in signal. Above this point, however, the limiter becomes overloaded, and further large increases in signal will not give any increase in output. To operate successfully, the limiter must be supplied with a large amount of signal, so that the amplitude of its output will not change for rather wide variations in amplitude of the signal. Noise, which causes little frequency modulation but much amplitude modulation of the received signal, is virtually wiped out in the limiter. The voltage across the grid resistor varies with the amplitude of the received signal. For this reason, conventional amplitude modulated signals may be received on the FM receiver by connecting the input of the audio amplifier to the top of this resistor, rather than to the discriminator output. When properly filtered. www.americanradiohistory.com HANDBOOK by a simple R -C circuit, the voltage across the grid resistor may also be used as a -v -c voltage for the receiver. When the limiter is FROM DISCRIMINATOR One of the 220 R2100 , C'C. K, 321 TO AUDIO GRID ME'. T R' most important factors in the design of an FM receiver is the frequency swing which it is intended to handle. It will be apparent from figure 17 that if the straight portion of the discriminator circuit covers a wider range of frequencies than those generated by the transmitter, the audio output will be reduced from the maximum value of which the receiver is capable. In this respect, the term "modulation percentage" is more applicable to the FM receiver than it is to the transmitter, since the modulation capability of the communication system is limited by the receiver bandwidth and the discriminator characteristic; full utilization of the linear portion of the characteristic amounts, in effect, to 100 per cent modulation. This means that some sort of standard must be agreed upon, for any particular type of communication, to make it unnecessary to vary the transmitter swing to accommodate different receivers. Two considerations influence the receiver bandwidth necessary for any particular type of communication. These are the maximum audio frequency which the system will handle, and the deviation ratio which will be employed. For voice communication, the maximum audio frequency is more or less fixed at 3000 to 4000 cycles. In the matter of deviation ratio, however, the amount of noise suppression which the FM system will provide is influenced by the ratio chosen, since the improvement in signal -to -noise ratio which the FM system shows over amplitude modulation is equivalent to a constant multiplied by the deviation ratio. This assumes that the signal is somewhat stronger than the noise at the receiver, however, as the advantages of wideband FM in regard to noise suppression disappear when the signal -to -noise ratio approaches unity. On the other hand, a low deviation ratio is more satisfactory for strictly communication work, where readability at low signal -to -noise ratios is more important than additional noise suppression when the signal is already appreciably stronger than the noise. As mentioned previously, broadcast FM practice is to use a deviation ratio of 5. When this ratio is applied to a voice -communication system, the total swing becomes 30 to 40 kc. With lower deviation ratios, such as are most frequently used for voice work, the swing becomes proportionally less, until at a deviation ratio of 1 the swing is equal to twice the highest audio frequency. Actually, however, the re- f C operating properly, a.v.c. is neither necessary nor desirable, however, for FM reception alone. Receiver Design Considerations Adapter NBFM L 340 uuF 730 uuF R'47It, C' IE00uur. R' 22 R, C' 3400 SWF. Figure 22 75- MICROSECOND DE- EMPHASIS CIRCUITS The audio signal transmitted by FM and TV stations has received high -frequency preemphasis, so that a de- emphasis c i r c u i t should be included between the output of the FM detector and the input of the audio system. ceiver bandwidth must be slightly greater than the expected transmitter swing, since for distortionless reception the receiver must pass the complete band of energy generated by the transmitter, and this band will always cover a range somewhat wider than the transmitter swing. Standards in FM broadcast and TV sound work call for the pre- emphasis of all audio modulating frequencies above about 2000 cycles, with a rising slope such as would be produced by a 75- microsecond RL network. Thus the FM receiver should include a compensating de- emphasis RC network with a time constant of 75 microseconds so that the overall frequency response from microphone to loudspeaker will approach linearity. The use of pre- emphasis and de- emphasis in this manner results in a considerable improvement in the overall signal -to -noise ratio of an FM system. Appropriate values for the de- emphasis network, for different values of circuit impedance are given in figure 22. Pre -Emphasis and De- Emphasis The unit diagrammed in figure 23 is designed to provide NBFM reception when attached to any communication receiver having a 455 -kc. i -f amplifier. Although NBFM can be received on an AM receiver by tuning the receiver to one side or the other of the incoming signal, a tremendous improvement in signal -to -noise ratio and in signal to amplitude ratio will be obtained by the use of a true FM detector system. The adapter uses two tubes. A 6AU6 is used as a limiter, and a 6AL5 as a discriminator. The audio level is approximately 10 A NBFM 455 -kc. Adapter Unit Radio Teletype Transmission FM 322 6AL5 6AÚ6 IOOK SODUF loo AUDIO OUT UL 100K ro /JUT 1D0(, 405 KC. I.F. IN 100 220K LUF VOLT- METER 0.1 W T 2 e+ 250 I -J.W. MILLEN 0i2-C3 V AT 3 MA. NOTE: ALL CAPACITORS IN /JP UNLESS OTHERWISE NOTED ALL RESISTORS 0.5 WATT UNLESS OTHERWISE NOTED Figure 23 NBFM ADAPTER FOR 455-KC. I -F SYSTEM volts peak for the maximum deviation which can be handled by a conventional 455 -kc. i -f system. The unit may be tuned by placing a high resistance d -c voltmeter across R, and tuning the trimmers of the i-f transformer for maximum voltage when an unmodulated signal is injected into the i -f strip of the receiver. The voltmeter should next be connected across the audio output terminal of the discriminator. The receiver is now tuned back and forth across the frequency of the incoming signal, and the movement of the voltmeter noted. When the receiver is exactly tuned on the signal the voltmeter reading should be zero. When the receiver is tuned to one side of center, the voltmeter reading should increase to a maximum value and then decrease gradually to zero as the signal is tuned out of the passband of the receiver. When the receiver is tuned to the other side of the signal the voltmeter should increase to the same maximum value but in the opposite direction or polarity, and then fall to zero as the signal is tuned out of the passband. It may be necessary to make small adjustments to C, and C, to make the voltmeter read zero when the signal is tuned in the center of the passband. 16 -5 Radio Teletype The teletype machine is an electric typewriter that is stimulated by d.c. pulses originated by the action of a second machine. The pulses may be transmitted from one machine to another by wire, or by a radio signal. When radio transmission is used, the system is termed radio teletype (RTTY). The d.c. pulses that comprise the teletype signal may be converted into three basic types of emission suitable for radio transmission. These are: 1- Frequency shift keying (FSK), designated as F1 emission; 2- Make -break keying (MBK),designated as Al emission. and; 3- Audio frequency shift keying (AFSK), designated as F2 emission. Frequency shift keying is obtained by varying the transmitted frequency of the radio signal a fixed amount (usually 850 cycles) during the keying process. The shift is accomplished in discrete intervals designated mark and space. Both types of intervals convey information to the teletype printer. Make -break keying is analogous to simple c -w transmission in that the radio carrier conveys information by changing from an off to an on condition. Early RTTY circuits employed MBK equipment, which is rapidly becoming obsolete since it is inferior to the frequency shift system. Audio frequency shift keying employs a steady radio carrier modulated by an audio tone that is shifted in frequency according to the RTTY pulses. Other forms of information transmission may be employed by a RTTY system which also encompass the translation of RTTY pulses into r -f signals. The RTTY code consists of the 26 letters of the alphabet, the space, the line feed, the carriage return, the bell, the upper case shift, and the lower case shift; making a total of 32 coded groups. Numerals, punctuation, and symbols may be taken care of in the case shift, since all transmitted letters are capitals. The FSK system normally employs the higher radio frequency as the mark, and the lower frequency as the space. This relationship holds true in the AFSK system also. The lower audio frequency (mark) is normally 2125 cycles and the higher audio tone (space) is 2975 cycles, giving a frequency difference of 850 Teletype Coding cycles. The Teletype System simple FSK teletype system may be added to any c -w transA mitter. The teletype keyboard prints the keyed letters on a tape, and at the same time generates the electrical code group that describes the letter. The d.c. pulses are impressed upon a distributor unit which arranges the typing and spacing pulses in proper sequence. The resulting series of impulses are applied to the transmitter frequency control device, which may be a reactance modulator, actuated by a polar relay. The received signal is hetrodyned against a beat oscillator to provide the two audio tones which are limited in amplitude and passed through audio filters to separate them. Rectification of the tones permits operation of a polar relay which can provide d.c. pulses suitable for operation of the tele- typewriter. www.americanradiohistory.com CHAPTER SEVENTEEN Sideband Transmission While single- sideband transmission (SSB) has attracted significant interest on amateur frequencies only in the past few years, the principles have been recognized and put to use in various commercial applications for many years. Expansion of single -sideband for both commercial and amateur communication has awaited the development of economical components possessing the required characteristics (such as sharp cutoff filters and high stability crystals) demanded by SSB techniques. The availability of such components and precision test equipment now makes possible the economical testing, adjustment and use of SSB equipment on a wider scale than before. Many of the seemingly insurmountable obstacles of past years no longer prevent the amateur from achieving the advantages of SSB for his class of operation. 17 -1 Commercial Applications of SSB Before discussion of amateur SSB equipment, it is helpful to review some of the commercial applications of SSB in an effort to avoid problems that are already solved. The first and only large scale use of SSB has been for multiplexing additional voice circuits on long distance telephone toll wires. Carrier systems came into wide use during the 30's, accompanied by the development of high Q toroids and copper oxide ring modulators of controlled characteristics. The problem solved by the carrier system was that of translating the 300 -3000 cycle voice band of frequencies to a higher frequency (for example, 40.3 to 43.0 kc.) for transmission on the toll wires, and then to reverse the translation process at the receiving terminal. It was possible in some short -haul equipment to amplitude modulate a 40 kilocycle carrier with the voice frequencies, in which case the resulting signal would occupy a band of frequencies between 37 and 43 kilocycles. Since the transmission properties of wires and cable deteriorate rapidly with increasing frequency, most systems required the bandwidth conservation characteristics of single -sideband transmission. In addition, the carrier wave was generally suppressed to reduce the power handling capability of the repeater amplifiers and diode modulators. A substantial body of literature on the components and circuit techniques of SSB has been generated by the large and continuing development effort to produce economical carrier telephone systems. The use of SSB for overseas radiotelephony has been practiced for several years though the number of such circuits has been numerically small. However, the economic value of such circuits has been great enough to warrant elaborate station equipment. It is from these stations that the impression has been obtained that SSB is too complicated for all but a corps of engineers and technicians to handle. Components such as lattice filters with 40 or more crystals have suggested astronomical expense. 323 www.americanradiohistory.com T H Sideband Transmission 324 e ,10lllllllllllu IIIIIIIIIIIII1III UPPER LOWER SIDEBAND I CARRIER SIDEDAND FRED. CARRIER ENVELOPE WITH COMBLE+ MODULATING WAVE FREQUENCY SPECTRUM WITH COMPLEX MODULATING WA, Figure REPRESENTATION OF A CONVENTIONAL AM SIGNAL 1 More recently, SSB techniques have been used to multiplex large numbers of voice channels on a microwave radio band using equipment principally developed for telephone carrier applications. It should be noted that all production equipment employed in these services uses the filter method of generating the single -sideband signal, though there is a wide variation in the types of filters actually used. The SSB signal is generated at a low frequency and at a low level, and then translated and linearly amplified to a high level at the operating frequency. Considerable development effort has been expended on high level phasing type transmitters wherein the problems of linear amplification are exchanged for the problems of accurately controlled phase shifts. Such equipment has featured automatic tuning circuits, servo- driven to facilitate frequency changing, but no transmitter of this type has been sufficiently attractive to warrant appreciable production. 17 -2 Derivation of Single -Sideband Signals The single -sideband method of communication is, essentially, a procedure for obtaining more efficient use of available frequency spectrum and of available transmitter capability. As a starting point for the discussion of single- sideband signals, let us take a conventional AM signal, such as shown in figure 1, as representing the most common method for transmitting complex intelligence such as voice or music. It will be noted in figure 1 that there are three distinct portions to the signal: the carrier, and the upper and the lower sideband group. These three portions always are present in a conventional AM signal. Of all these portions the carrier is the least necessary and the most expensive to transmit. It is an actual E R A D I O fact, and it can be proved mathematically (and physically with a highly selective receiver) that the carrier of an AM signal remains unchanged in amplitude, whether it is being modulated or not. Of course the carrier appears to be modulated when we observe the modulated signal on a receiving system or indicator which passes a sufficiently wide band that the carrier and the modulation sidebands are viewed at the same time. This apparent change in the amplitude of the carrier with modulation is simply the result of the sidebands beating with the carrier. However, if we receive the signal on a highly selective receiver, and if we modulate the carrier with a sine wave of 3000 to 5000 cycles, we will readily see that the carrier, or either of the sidebands can be tuned in separately; the carrier amplitude, as observed on a signal strength meter, will remain constant, while the amplitude of the sidebands will vary in direct proportion to the modulation percentage. Elimination of the Carrier and It is obvious from the pre vious discussion that the carrier is superfluous so far as the transmission of intelligence is concerned. It is obviously a convenience, however, since it provides a signal at the receiving end for the sidebands to beat with and thus to reproduce the original modulating signal. It is equally true that the transmission of both sidebands under ordinary conditions is superfluous since identically the same intelligence is contained in both side bands. Several systems for carrier and sideband elimination will be discussed in this chapter. One Sideband Power Advantage SSB over AM Single sideband is a very efficient form of voice communication by radio. The amount of radio frequency spectrum occupied can be no greater than the frequency range of the audio or speech signal transmitted, whereas other forms of radio transmission require from two to several times as much spectrum space. The r -f power in the transmitted SSB signal is directly proportional to the power in the original audio signal and no strong carrier is transmitted. Except for a weak pilot carrier present in some commercial usage, there is no r -f output when there is no audio input. The power output rating of a SSB transmitter is given in terms of peak envelope pcwer (PEP) . This may be defined as the r -m -s power at the crest of the modulation of www.americanradiohistory.com HANDBOOK Derivation envelope. The peak envelope power of a conventional amplitude modulated signal at 100% modulation is four times the carrier power. The average power input to a SSB transmitter is therefore a very small fraction of the power input to a conventional amplitude modulated transmitter of the same power rating. Single sideband is well suited for long range communications because of its spectrum and power economy and because it is less susceptible to the effects of selective fading and interference than amplitude modulation. The principal advantages of SSB arise from the elimination of the high- energy carrier and from further reduction in sideband power permitted by the improved performance of SSB under unfavorable propagation conditions. In the presence of narrow band man -made interference, the narrower bandwidth of SSB reduces the probability of destructive interference .A statistical study of the distribution of signals on the air versus the signal strength shows that the probability of successful communication will be the same if the SSB power is equal to one -half the power of one of the two a -m sidebands. Thus SSB can give from 0 to 9 db improvement under various conditions when the total sideband power is equal in SSB and a -m. In general, it may be assumed that 3 db of the possible 9 db advantage will be realized on the average contact. In this case, the SSB -power required for equivalent performance is equal to the power in one of the a -m sidebands. For example, this would rate a 100 -watt SSB and a 400 watt (carrier) a -m transmitter as having equal performance. It should be noted that in this comparison it is assumed that the receiver bandwidth is just sufficient to accept the transmitted intelligence in each case. To help evaluate other methods of comparison the following points should be considered. In conventional amplitude modulation two sidebands are transmitted, each having a peak envelope power equal to 1/4-carrier power. For example, a 100 -watt a -m signal will have 25watt peak envelope power in each sideband, or a total of 50 watts. When the receiver detects this signal, the voltages of the two side bands are added in the detector. Thus the detector output voltage is equivalent to that of a 100 -watt SSB signal. This method of comparison says that a 100 watt SSB transmitter is just equivalent to a 100 -watt a -m transmitter. This assumption is valid only when the receiver bandwidth used for SSB is the same as that required for amplitude modulation KC. KC AUDIO SPECTRUM 4004RC' SSB SPECTRUM (UPPER SIOEIAMO) `3996 KC 325 4000 KC SSB SPECTRUM (LOWER S/OEBAAO ) Figure 2 RELATIONSHIP OF AUDIO AND SSB SPECTRUMS The single sideband components are the same the original the frequency quency of the of the various as audio components except that of each is raised by the frecarrier. The relative amplitude components remains the same. (e.g., 6 kilocycles) , when there is no noise or interference other than broadband noise, and if the a -m signal is not degraded by propagation. By using half the bandwidth for SSB reception ( e.g., 3 kilocycles) the noise is reduced 3 db so the 100 watt SSB signal becomes equivalent to a 200 watt carrier a -m signal. It is also possible for the a -m signal to be degraded another 3 db on the average due to narrow band interference and poor propagation conditions, giving a possible 4 to 1 power advantage to the SSB signal. It should be noted that 3 db signal -to -noise ratio is lost when receiving only one sideband of an a -m signal. The narrower receiving bandwidth reduces the noise by 3 db but the 6 db advantage of coherent detection is lost, leaving a net loss of 3 db. Poor propagation will degrade this "one sideband" reception of an a -m signal less than double sideband reception, however. Also under severe narrow band interference conditions (e.g., an adjacent strong signal) the ability to reject all interference on one side of the carrier is a great advantage. The Nature of SSB Signal The nature of a single sideband signal is easily visualized by noting that the SSB signal components are exactly the same as the original audio components except that the frequency of each is raised by the frequency of the carrier. The relative amplitude of the various components remains the same, however. (The first statement is only true for the upper sideband since the lower sideband frequency components are the difference between the carrier and the original audio signal). Figure 2A, B, and C shows how the audio spectrum is simply moved up into the radio spectrum to give the upper sideband. The lower sideband is the same except inverted, as shown in figure 2C. Either sideband may be used. It is apparent that the carrier frequency www.americanradiohistory.com o THE RADIO Sideband Transmission 326 SINGLE TONE Figure 3 A SINGLE SINE WAVE TONE INPUT TO A SSB TRANSMITTER RESULTS IN A STEADY SINGLE SINE WAVE R -F OUTFIT (A). TWO AUDIO TONES OF EQUAL AMPLITUDE BEAT TOGETHER TO PRODUCE HALF -SINE WAVES AS SHOWN IN (B). of a SSB signal can only be changed by adding or subtracting to the original carrier frequency. This is done by heterodyning, using converter or mixer circuits similar to those employed in a superheterodyne receiver. It is noted that a single sine wave tone input to a SSB transmitter results in a single steady sine wave r -f ouput, as shown in figure 3A. Since it is difficult to measure the performance of a linear amplifier with a single tone, it has become standard practice to use two tones of equal amplitude for test purposes. The two radio frequencies thus pro duced beat together to give the SSB envelope shown in figure 3B. This figure has the shape of half sine waves, and from one null to the next represents one full cycle of the difference frequency. How this envelope is generated is shown more fully in figures 4A and 4B. f, and f2 represent the two tone signals. When a vector representing the lower frequency tone signal is used as a reference, the other vector rotates around it as shown, and this action Figure 5 TWO -TONE SSB ENVELOPE WHEN ONE TONE HAS TWICE THE AMPLITUDE OF THE OTHER. Figure 6 THREE -TONE SSB ENVELOPE WHEN EQUAL TONES OF EQUAL FREQUENCY SPACINGS ARE USED. generates the SSB envelope When the two vectors are exactly opposite in phase, the output is zero and this causes the null in the envelope. If one tone has twice the amplitude of the other, the envelope shape is shown in figure 5. Figure 6 shows the SSB envelope of three equal tones of equal frequency spacings and at one particular phase relationship. Figure 7A shows the SSB envelope of four equal tones with equal frequency spacings and at one particular phase relationship. The phase relationships chosen are such that at some instant the vectors representing the several tones are all in phase. Figure 7B shows a SSB envelope of a square wave. A pure square wave requires infinite bandwidth, so its SSB envelope requires infinite amplitude. This emphasizes the point that the SSB envelope shape is not the same as the original audio wave shape, and usually bears no similarity to it. This is because the percentage difference between the radio frequencies is very small, even though one audio tone may be several times the other in terms of frequency. Speech clipping as used r, FREQUENCY fi © 12 OF CRRiER Figure 7A FOUR TONE SSB ENVELOPE Figure 4 VECTOR REPRESENTATION OF TWO -TONE SSB ENVELOPE when equal tones with equal frequency spacings are used www.americanradiohistory.com Figure 7B SSB ENVELOPE OF A SQUARE WAVE. Peak of wave reaches infinite amplitude. HANDBOOK Derivation in amplitude modulation is of no practical value in SSB because the SSB r -f envelopes are so different than the audio envelopes. A heavily clipped wave approaches a square wave and a square wave gives a SSB envelope with peaks of infinite amplitude as shown in figure 7B. Carrier Frequency Stability Requirements o R G T PUSH -PULL AUDIO IN Reception of a SSB signal is accomplished by simply heterodyning the carrier down to zero frequency. (The conversion frequency used in the last heterodyne step is often called the reinserted carrier). If the SSB signal is not heterodyned down to exactly zero frequency, each frequency component of the detected audio signal will be high or low by the amount of this error. An error of 10 to 20 c.p s. for speech signals is acceptable from an intelligibility standpoint, but an error of the order of 50 c.p.s. seriously degrades the intelligibility. An error of 20 c.p.s. is not acceptable for the transmission of music, however, because the harmonic relationship of the notes would be destroyed. For example, the harmonics of 220 c.p.s. are 440, 660, 880, etc., but a 10 c.p s. error gives 230, 450, 670, 890, etc., or 210, 430, 650, 870, etc., if the original error is on the other side. This error would destroy the original sound of the tones, and the harmony between the tones. Suppression of the carrier is common in amateur SSB work, so the combined frequency stabilities of all oscillators in both the transmitting and receiving equipment add together to give the frequency error found in detection. In order to overcome much of the frequency stability problem, it is common commercial practice to transmit a pilot carrier at a reduced amplitude. This is usually 20 db below one tone of a two -tone signal, or 26 db below the peak envelope power rating of the transmitter. This pilot carrier is filtered out from the other signals at the receiver and either amplified and used for the reinserted carrier or used to control the frequency of a local oscillator. By this means, the frequency drift of the carrier is eliminated as an error in detection. Advantage of SSB with Selective Fading 327 On long distance corn - munication using a -m, circuits selective fading often causes severe distortion and at times makes the signal unintelligible. When one sideband is weaker than the other, distor- R r OUT 0 Figure 8 SHOWING TWO COMMON TYPES OF BALANCED MODULATORS that o balanced modulator changes the circuit condition from single ended to push -pull, or vice versa. Choice of circuit depends upon external circuit conditions since both the (A) and B: arrangements can give Notice satisfactory generation of a double -sideband suppressed- carrier signal. tien results; but when the carrier becomes weak and the sidebands are strong, the distortion is extremely severe and the signal may sound like "monkey chatter." This is because a carrier of at least twice the amplitude of either sideband is necessary to demodulate the signal properly. This can be overcome by using exalted carrier reception in which the carrier is amplified separately and then reinserted before the signal is demodulated or detected. This is a great help, but the reinserted carrier must be very close to the same phase as the original carrier. For example, if the reinserted carrier were 90 degrees from the original source, the a -m signal would be converted to phase modulation and the usual a -m detector would deliver no output. The phase of the reinserted carrier is of no importance in SSB reception and by using a strong reinserted carrier, exalted carrier reception is in effect realized. Selective fading with one sideband simply changes the amplitude and the frequency response of the system and very seldom causes the signal to become unintelligible. Thus the receiving techniques used with SSB are those which inherently greatly minimize distortion due to selective fading. www.americanradiohistory.com 328 THE RADIO Sideband Transmission MO slDE- DAN: 22 VOLTAGE. -,) z, OUTPUT SHUNT-QUAD MODULATOR BRIDGE MODULATOR CARRIER vOLTAGE SiDE- 7 BAND OUTPUT z, RING MODULATOR - DOUBLE-BALANCED MODULATOR GAI ggiER I VOLTAGE Figure 9 TWO TYPES OF DIODE BALANCED MODULATOR balanced modulator circuits are commonly used in carrier telephone work and in single-sideband systems where the carrier frequency and modulating frequency are relatively close together. Vacuum diodes, copper oxide rectifiers, or crystal diodes may be used in the circuits. Such 17 -3 Carrier Elimination Circuits Various circuits may be employed to eliminate the carrier to provide a double sideband signal. A selective filter may follow the carrier elimination circuit to produce a single sideband signal. Two modulated amplifiers may be connected with the carrier inputs 180° out of phase, and with the carrier outputs in parallel. The car- rier will be balanced out of the output circuit, leaving only the two sidebands. Such a circuit is called a balanced modulator. Any non -linear element will produce modulation. That is, if two signals are put in, sum and difference frequencies as well as the original frequencies appear in the output. This phenomenon is objectionable in amplifiers and desirable in modulators or mixers. In addition to the sum and difference frequencies, other outputs (such as twice one frequency plus the other) may appear. All combinations of all harmonics of each input frequency may appear, but in general these are of decreasing amplitude with increasing order of harmonic. These outputs are usually rejected by selective circuits following the modulator. All modulators are not alike in the magnitude of these higher order outputs. Balanced diode rings operating in the square law region are fairly good and pentagrid converters much poorer. Excessive carrier level in tube mixers will increase the relative magnitude of the higher order outputs. Two types of triode balanced modulators are shown in figure 8, and two types of diode modulators in figure 9. Balanced modulators employing vacuum tubes may be made to work very easily to a point. Circuits may be devised wherein both input signals may be applied to a high impedance grid, simplifying isolation and loading problems. The most important difficulties with these vacuum tube modulator circuits are: (1) Balance is not independent of signal level. (2) Balance drifts with time and environment. (3) The carrier level for low "high order output" is critical, and (4) Such circuits have limited dynamic range. A number of typical circuits are shown in figure 10. Of the group the most satisfactory performance is to be had from plate modulated triodes. IDO o-.1 0.I O T R.r CARIRR IN PULL AUDIO IN PU SII PLATE MODULATED BALANCED TRIODE MODULATOR BALANCED TRIODE MODULATOR WITH SINGLE ENDED INPUT CIRCUITS Figure 10 BALANCED MODULATORS www.americanradiohistory.com BALANCED PENTAGRID CONVERTER MODULATOR HANDBOOK MODULA T. VOLTAGE Carrier Elimination SIDEBAND OurPUr CARRIER VOLTAGE HIGH Z MODULATING VOLTAGE HIGH Z SIDEBAND OUTPUT LOW Z LOW Z MODULATING VOLTAGE CARRIER VOLTAGE DOUBLE- BALANCED RING MODULATOR 329 SIDEBAND OUTPUT CARRIER VOLTAGE SHUN.' QUAD MODAL ATOR SERIES -QUAD MODULATOR Figure 11 DIODE RING MODULATORS Modulation in telephone carrier equipment has been very successfully accomplished with copper -oxide double balanced ring modulators. More recently, germanium diodes have been applied to similar circuits. The basic diode ring circuits are shown in figure 11. The most widely applied is the double balanced ring (A) . Both carrier and input are balanced with respect to the output, which is advantageous when the output frequency is not sufficiently different from the inputs to allow ready separation by filters. It should be noted that the carrier must pass through the balanced input and output transformers. Care must be taken in adapting this circuit to minimize the carrier power that will be lost in these elements. The shunt and series quad circuits are usable when the output frequencies are entirely different (i.e.: audio and r.f.) The shunt quad (B) is used with high source and load impedances and the series quad (C) with low source and load impedances. These two circuits may be adapted to use only two diodes, substituting a balanced transformer for one side of the bridge, as shown in figure 12. It should be noted that these circuits present a half -wave load to the carrier source. In applying any of these circuits, r -f chokes and capacitors must be employed to control the path of signal and carrier currents. In the shunt pair, for example, a blocking capacitor is used to prevent the r -f load from shorting the audio input. To a first approximation, the source and load impedances should be an arithmetical mean of the forward and back resistances of the diodes employed. A workable rule of thumb is that the source and load impedances be ten to twenty times the forward resistance for semi -conductor rings. The high frequency limit of operation in the case of junction and copper-oxide diodes may be appreciably extended by the use of very low source and load impedances. Copper -oxide diodes suitable for carrier Diode Ring Modulators . work are normally manufactured to order. They offer no particular advantage to the amateur, though their excellent long -term stability is important in commercial applications. Rectifier types intended to be used as meter rectifiers are not likely to have the balance or high frequency response desirable in amateur SSB transmitters. Vacuum diodes such as the 6AL5 may be used as modulators. Balancing the heater cathode capacity is a major difficulty except when the 6AL5 is used at low source and load impedance levels. In addition, contact potentials of the order of a few tenths of a volt may also disturb low level applications (figure 13). The double diode circuits appear attractive, but in general it is more difficult to balance a transformer at carrier frequency than an additional pair of diodes. Balancing potentiometers may be employed, but the actual cause of the unbalance is far more subtile, and cannot be adequately corrected with a single adjustment. A signal produced by any of the above circuits may be classified as a double sideband, suppressed- carrier signal. MODULATING VOLTAGE I_ SIDEBAND OUTPUT SHUNT -PAIR MODULATOR CARRIER VOLTAGE O SERIES -PAIR MODULATOR MODULATI NC VOLTAGE ARRIER VOLTAGE Figure 12 DOUBLE -DIODE PAIRED MODULATORS www.americanradiohistory.com THE RADIO Sideband Transmission 330 B i PUSH-PULL R CARRIER IN F R.F OUT O OA SERIES- BALANCED DIODE MODULATOR USING 6AL5 TUBE 6AL5 -6 -5 -4 -3 -2 -1 0 KILOCYCLES DEVIATION R.F.CAR RIES IN Figure 15 BANDPASS CHARACTERISTIC OF BURNELL S -15000 SINGLE SIDEBAND FILTER R.F. OUT oRFC AUDIO IN 001 OB RING -DIODE MODULATOR USING 6AL5 TUBE Figure 13 VACUUM DIODE MODULATOR CIRCUITS Generation of Single -Sideband Signals 17 -4 In general, there are two commonly used methods by which a single -sideband signal may be generated. These systems are: (I) The Filter Method, and (2) The Phasing Method. The systems may be used singly or in combination, and either method, in theory, may be used at the operating frequency of the transmitter or at some other frequency with the signal at the operating frequency being obtained through the use of frequency changers (mixers) . The Filter Method The filter method for obtaining signal is the classic method which has been in use by the telephone companies for many years both for a SSB 100-10000 SPE ECM AMPLIFIER I7-SO ZOO-50001. '1. SPEECH FILTER 50-53 47-SO KC. NC 47 -50 KC SIDEBAND FILTER BALANCED MODULATOR land -line and radio communications. The mode of operation of the filter method is diagrammed in figure 14, in terms of components and filters which normally would be available to the amateur or experimenter. The output of the speech amplifier passes through a conventional speech filter to limit the frequency range of the speech to about 200 to 3000 cycles. This signal then is fed to a balanced modulator along with a 50,000 -cycle first carrier from a self-excited oscillator. A low -frequency balanced modulator of this type most conveniently may be made up of four diodes of the vacuum or crystal type cross connected in a balanced bridge or ring modulator circuit. Such a modulator passes only the sideband components resulting from the sum and difference between the two signals being fed to the balanced modulator. The audio signal and the 50 -kc. carrier signal from the oscillator both cancel out in the balanced modulator so that a band of frequencies between 47 and 50 kc. and another band of frequencies between 50 and 53 kc. appear in the output. The signals from the first balanced modulator are then fed through the most critical 47-5014C PHASE INVERTER KC SO KC. OSCILLATOR BLOCK BALANCED MODULATOR 1750-1950 KC. OSCILLATOR Figure 14 DIAGRAM OF FILTER EXCITER EMPLOYING A 50 -K.C. SIDEBAND FILTER www.americanradiohistory.com HIGH -O TUNEDCIRCUIT FOR OUTPUT IN 1100 - 2000 AC. SAND HANDBOOK Generation of S.S.B. 6AL5 2 12AU7 6AU6 SNUNTDIODE MODULATOR 15MH 250LU1F 01 ( SO i 331 6C4 12AU7 RF AMPLIFIER AMASE-INVERTER .01 33111f LO RC.) G P MIC SO KCR. FILTE B 002-- { 0.35 r 005 I. +350 V SOMA -PULL R.F. TO BALANCED MODULATOR FOR CONVERSION TO 160 METERS PUSH 12AU7 NOTE OSCILLATOR SO KC : UNLESS OTHERWISE SPECIFIED, RESISTORS ARE 0.5 WATT. CAPACITORS /AI 4/F. 00A 100 Figure 16 OPERATIONAL CIRCUIT FOR SSB EXCITER USING THE BURNELL 50 -KC. SIDEBAND FILTER component in the whole system -the first sideband filter. It is the function of this first sideband filter to separate the desired 47 to 50 kc. sideband from the unneeded and undesired 50 to 53 kc. sideband. Hence this filter must have low attenuation in the region between 47 and 50 kc., a very rapid slope in the vicinity of 50 kc., and a very high attenuation to the sideband components falling between 50 and 53 kilocycles. Burnell & Co., Inc., of Yonkers, New York produce such a filter, designated as Burnell S -1 5,000. The passband of this filter is shown in figure 15. Appearing, then, at the output of the filter is a single sideband of 47 kc. to 50 kc. This sideband may be passed through a phase inverter to obtain a balanced output, and then fed to a balanced mixer. A local oscillator operating in the range of 1750 kc. to 1950 kc. is used as the conversion oscillator. Additional conversion stages may now be added to trans200-30001 SPEECH AMPLIFIER 111111 200-3000 LOW 2 PHASE INVERTER 453 SHUNT -QUAD RING MODULATOR late the SSB signal to the desired frequency. Since only linear amplification may be used, it is not possible to use frequency multiplying stages. Any frequency changing must be done by the beating-oscillator technique. An operational circuit of this type of SSB exciter is shown in figure 16. A second type of filter-exciter for SSB may be built around the Collins Mechanical Filter. Such an exciter is diagrammed in figure 17. Voice frequencies in the range of 200 -3000 cycles are amplified and fed to a low impedance phase -inverter to furnish balanced audio. This audio, together with a suitably chosen r -f signal, is mixed in a ring modulator, made up of small germanium diodes. Depending upon the choice of frequency of the r -f oscillator, either the upper or lower sideband may be applied to the input of the mechanical filter. The carrier, to some extent, has been rejected by the ring modulator. Additional carrier rejection is afforded by the excellent passband -158 RC. - 453-458 RC. 455 MECHANICAL F 3953 RC. K C. ILTER CONVERTER FOR OUTPUT ON 3953 R 450-453 KC. 453 K.C. OSCILLATOR 3500 K.C. OSCILLATOR Figure 17 BLOCK DIAGRAM OF FILTER EXCITER EMPLOYING A 455 -KC. MECHANICAL FILTER FOR SIDEBAND SELECTION www.americanradiohistory.com R.F. AMPLIFIER WITH HIGH -Q TUNED CIRCUIT C. THE RADIO Sideband Transmission 332 CMSO CMSO 0 10 20 CMSO CM SO fT -EII CHANNEL AT-241 CHANNEL If CRYSTAL SO CRYSTAL 4II. J NC. .- FREQUENCY CARRIER 465.PMC. Figure 18 SIMPLE CRYSTAL LATTICE FILTER < e0 70 459 480 461 462 463 464 FREQUENCY (KC.) characteristics of the mechanical filter. For simplicity, the mixing and filtering operation usually takes place at a frequency of 455 kilocycles. The single -sideband signal appearing at the output of the mechanical filter may be translated directly to a higher operating frequency. Suitable tuned circuits must follow the conversion stage to eliminate the signal from the conversion oscillator. The heart of a filter -type SSB exciter is the sideband filter. Conventional coils and capacitors may be used to construct a filter based upon standard wave filter techniques. The Q of the filter inductances must be high when compared with the reciprocal of the fractional bandwidth. If a bandwidth of 3 kc. is needed at a carrier frequency of 50 kc., the bandwidth expressed in terms of the carrier frequency is 3/50 or 6%. This is expressed in terms of fractional bandwidth as 1/16. For satisfactory operation, the Wave Filters o I 10 UPPER SI MOAN 7WE CARRIER FREQUENCY 30 60 246 247 248 249 250 251 252 253 254 FREQUENCY (K.C.) Figure 19 PASSBAND OF LOWER AND UPPER SIDEBAND MECHANICAL FILTER 235 Q of the filter inductances should be 10 times the reciprocal of this, or 160. Appropriate Q is generally obtained from toroidal inductances, though there is some possibility of using iron core solenoids between 10 kc. and 20 kc. A characteristic impedance below 1000 ohms should be selected to prevent distributed capacity of the inductances from spoiling overall performance. Paper capacitors intended for bypass work may not be trusted for stability or low loss and should not be used in filter circuits. Care should be taken that the levels of both accepted and rejected signals are low enough so that saturation of the filter inductances does not occur. The best known filter responses have been obtained with crystal filters. Types designed for program carrier service cut -off 80 db in less than 50 cycles. More than 80 crystals are used in this type of filter. The crystals are cut to control reactance and resistance as well as the resonant frequency. The circuits used are based on full lattices. The war -surplus low frequency crystals may be adapted to this type of filter with some success. Experimental designs usually synthesize a selectivity curve by grouping sharp notches at the side of the passband. Where the width of the passband is greater than twice the spacing of the series and parallel resonance of the crystals, special circuit techniques must be used. A typical crystal filter using these surplus crystals, and its approximate passband is shown in figure 18. Crystal Filters Filters using mechanical resonators ha v e been studied by a number of companies and are offered commercially by the Collins Radio Co. They are available in a variety of bandwidths Mechanical Filters www.americanradiohistory.com HANDBOOK Generation of S.S.B. 333 BALANCED SPC AMPLIFIEEEH R SPEECH FILTER MODULATO N I AUDIO PHASE 20Q JOOD OR NETWORKS Q2 AND 15=110* PHASE DIFFERENCE BETWEEN O/ PHASE DIFFERENCE BETWEEN RI AND RI = PO. AMPLIFIER STAGES DIRECTLY TO ANTENNA SYSTEM TO POWER SPLITTING BALANCED MODULATOF N. 2 ei 92 RADIO FRED. PHASE SPLITTING NETWORK RADIO FRED. SIGNAL AT CARRIER FRED. Figure 20 BLOCK DIAGRAM OF THE "PHASING" METHOD The phasing method of obtaining a single -sideband signal is simpler thon the filter system in regard to the number of tubes and circuits required. The system is also less expensive in regard to the components required, but is more critical in regard to adjustments for the transmission of a pure single -sideband signal. at center frequencies of 250 kc. and 455 kc. The 250 kc. series is specifically intended for sideband selection. The selectivity attained by these filters is intermediate between good LC filters at low center frequencies and engineered quartz crystal filters. A passband of two 250 kc. filters is shown in figure 19. In application of the mechanical filters some special precautions are necessary. The driving and pick -up coils should be carefully resonated to the operating frequency. If circuit capacities are unknown, trimmer capacitors should be used across the coils. Maladjustment of these tuned coils will increase insertion loss and the peak to- valley ratio. On high impedance filters ( ten to twenty thousand ohms) signals greater than 2 volts at the input should be avoided. D -c should be blocked out of the end coils. While the filters are rated for 5 ma. of coil current, they are not rated for d -c plate voltage. a number of points of view from which the operation of the phasing system of SSB generation may be described. We may state that we generate two double-sideband suppressed carrier signals, each in its own balanced modulator, that both the r -f phase and the audio phase of the two signals differ by 90 degrees, and that the outputs of the two balanced modulators are added with the result that one sideband is increased in amplitude and the other one is cancelled. This, of course, is a true description of the action that takes place. But it is much easier to consider the phasing system as a method simply of adding (or of subtracting) the desired modulation frequency and the nominal carrier frequency. The carrier frequency of course is not trans- The Phasing System There are mitted, as is the case with all SSB transmissions, but only the sum or the difference of the modulation band from the nominal carrier is transmitted ( figure 20 ) . The phasing system has the obvious advantage that all the electrical circuits which give rise to the single sideband can operate in a practical transmitter at the nominal output frequency of the transmiter. That is to say that if we desire to produce a single sideband whose nominal carrier frequency is 3.9 Mc., the balanced modulators are fed with a 3.9 -Mc. signal and with the audio signal from the phase splitters. It is not necessary to go through several frequency conversions in order to obtain a sideband at the desired output frequency, as in the case with the filter method of sideband generation. Assuming that we feed a speech signal to the balanced modulators along with the 3900 kc. carrier (3.9 Mc.) we will obtain in the output of the balanced modulators a signal which is either the sum of the carrier signal and the speech band, or the difference between the carrier and the speech band. Thus if our speech signal covers the band from 200 to 3000 cycles, we will obtain in the output a band of frequencies from 3900.2 to 3903 kc. (the sum of the two, or the "upper" sideband), or a band from 3897 to 3899.8 kc. (the difference between the two or the "lower" sideband) . A further advantage of the phasing system of sideband generation is the fact that it is a very simple matter to select either the upper sideband or the lower sideband for transmission. A simple double -pole double -throw reversing switch in two of the four audio leads to the balanced modulators is all that is required. www.americanradiohistory.com Sideband Transmission 334 T H E R A D I O of a few milliwatts it is most common to make the first stage in the amplifier chain a class A amplifier, then to use one or more class B linear amplifiers to bring the output up to the desired level. R.F OUT o \ 180690270/ FOUR-PHASE A F. INDUCTIVE COUPLING 0 IBC' ar 270 FOUR -PHASE A. F. Figure 21 TWO CIRCUITS FOR SINGLE SIDEBAND GENERATION BY THE PHASING METHOD. The circuit of (A) offers the advantages of simplicity in the single-ended input circuits plus a push -pull output circuit. Circuit (8) requires double -ended input circuits but allows all the plates to be connected in parallel for the output circuit. High -Level Phasing vs. Low -Level Phasing Balanced Illustrated in figure 8 are the two basic balanced modulator circuits which give good results with a radio frequency carrier and an audio modulating signal. Note that one push -pull and one single ended tank circuit is required, but that the push -pull circuit may be placed either in the plate or the grid circuit. Also, the audio modulating voltage always is fed into the stage in push -pull, and the tubes normally are operated Class A. When combining two balanced modulators to make up a double balanced modulator as used in the generation of an SSB signal by the phasing system, only one plate circuit is required for the two balanced modulators. However, separate grid circuits are required since the grid circuits of the two balanced modulators operate at an r -f phase difference of 90 degrees. Shown in figure 21 are the two types of double balanced modulator circuits used for generation of an SSB signal. Note that the circuit of figure 21A is derived from the balanced modulator of figure 8A, and similarly figure 21B is derived from figure 8B. Another circuit that gives excellent performance and is very easy to adjust is shown in figure 22. The adjustments for carrier balance are made by adjusting the potentiometer for voltage balance and then the small variable capacitor for exact phase balance of the balanced carrier voltage feeding the diode modulator. Modulator Circuits MECHANICAL FILTER The plate -circuit efficiency of the four tubes usually used to make up the two balanced modulators of the phasing system may run as high as 50 to 70 per cent, depending upon the operating angle of plate current flow. Hence it is possible to operate the double balanced modulator directly into the antenna system as the output stage of the transmitter. The alternative arrangement is to generate the SSB signal at a lower level and then to amplify this signal to the level desired by means of class A or class B r -f power amplifiers. If the SSB signal is generated at a level VOLT SSB 0 1 OUTPUT R-F CARRIER 2.3 VOLTS BALANCED Figure 22 MODULATOR FOR USE WITH MECHANICAL FILTER www.americanradiohistory.com HANDBOOK 335 Generation of S.S.B. fISO V. 3900 .024220I1 TO BAL. MOO. I AUDIO SIGNAL 3900 3900 TO SAL 00l3! 9100 .2 20 K 7 B .150 Figure 23 LOW -Q R -F PHASE -SHIFT NETWORK The r -f phase -shift system illustrated above is convenient in a case where it is desired to make small changes in the operating frequency of the system without the necessity of being precise in the adjustment of two coupled circuits as used for r -f phase shift in the circuit of figure 21. Radio-Frequency Phasing A single -sideband genera - tor of the phasing type requires that the two balanced modulators be fed with r -f signals having a 90- degree phase difference. This r -f phase difference may be obtained through the use of two loosely coupled resonant circuits, such as illustrated in figures 21A and 21B. The r -f signal is coupled directly or inductively to one of the tuned circuits, and the coupling between the two circuits is varied until, at resonance of both circuits, the r -f voltages developed across each circuit have the same amplitude and a 90- degree phase difference. The 90- degree r -f phase difference also may be obtained through the use of a low -Q phase shifting network, such as illustrated in figure 23; or it may be obtained through the use of a lumped -constant quarter -wave line. The low Q phase- shifting system has proved quite practicable for use in single -sideband systems, particularly on the lower frequencies. In such an arrangement the two resistances R have the same value, usually in the range between 100 and a few thousand ohms. Capacitor C, in shunt with the input capacitances of the tubes and circuit capacitances, has a reactance at the operating frequency equal to the value of the resistor R. Also, inductor L has a net inductive reactance equal in value at the operating frequency to resistance R. The inductance chosen for use at L must take into account the cancelling effect of the input capacitance of the tubes and the circuit capacitance; hence the inductance should be v. Figure 24 DOME AUDIO -PHASE -SHIFT NETWORK This circuit arrangement is convenient for obtaining the audio phase shift when it is desired to use a minimum of circuit components and tube elements. variable and should have a lower value of inductance than that value of inductance which would have the same reactance as resistor R. Inductor L may be considered as being made up of two values of inductance in parallel; (a) a value of inductance which will resonate at the operating frequency with the circuit and tube capacitances, and ( b) the value of inductance which is equal in reactance to the resistance R. In a network such as shown in figure 23, equal and opposite 45- degree phase shifts are provided by the RL and RC circuits, thus providing a 90- degree phase difference between the excitation voltages applied to the two balanced modulators. Audio -Frequency The audio -frequency phaseshifting networks used in generating a single -sideband signal by the phasing method usually are based on those described by Dome in an article in the December, 1946, Electronics. A relatively simple network for accomplishing the 90- degree phase shift over the range from 160 to 3500 cycles is illustrated in figure 24. The values of resistance and capacitance must be carefully checked to insure minimum deviation from a 90- degree phase shift over the 200 to 3000 cycle range. Another version of the Dome network is shown in figure 25. This network employs Phasing three 12AU7 tubes and provides balanced output for the two balanced modulators. As with the previous network, values of the resistances within the network must be held to very close tolerances. It is necessary to restrict the speech range to 300 to 3000 cycles with this network. Audio frequencies outside this range will not have the necessary phase -shift at the output www.americanradiohistory.com 336 THE RADIO Sideband Transmission 12AÚ7 12AU7 12AU7 +105 V. REGULATED TO BAL. MOD Y 0.5 I R .01 PUSH -PULL AUDIO INPUT AUD INPUT N K 0.5 M0D2 B 2450 VLF Ill 607 Figure 26 11M PASSIVE AUDIO -PHASE-SHIFT NETWORK, USEFUL OVER RANGE OF 300 TO 3000 CYCLES. 30 1% +105 V. REGULATED Figure 25 A VERSION OF THE DOME AUDIO -PHASE-SHIFT NETWORK of the network and will show up as spurious emissions on the sideband signal, and also in the region of the rejected sideband. A low pass 3500 cycle speech filter, such as the Chicago Transformer Co. LPF -2 should be used ahead of this phase -shift network. A passive audio phase -shift network that employs no tubes is shown in figure 26. This network has the same type of operating restrictions as those described above. Additional information concerning phase -shift networks will be found in Single Sideband Techniques published by the Cowan Publishing Corp., New York, and The Single Sideband Digest published by the American Radio Relay League. A comprehensive sideband review is contained in the December, 1956 issue of Proceedings of the I.R.E. Comparison of Filter and Phasing Methods of SSB Generation 1!s MODSI TO SAL. 343 2 TO BAL MOD II 2 1)]]B TO SAL. Either the filter or the phasing method of single -sideband generation is theoretically capable of a high degree of performance. In general, it may be said that a high degree of unwanted signal rejection may be attained with less expense and circuit complexity with the filter method. The selective circuits for rejection of unwanted frequencies operate at a relativly low frequency, are designed for this one frequency and have a relatively high order of Q. Carrier rejection of the order of 50 db or so may be obtained with a relatively simple filter and a balanced modulator, and unwanted sideband rejection in the region of 60 db is economically possible. The phasing method of SSB generation exchanges the problems of high -Q circuits and linear amplification for the problems of accurately controlled phase -shift networks. If the phasing method is employed on the actual transmitting frequency, change of frequency must be accompanied by a corresponding rebalance of the phasing networks. In addition, it is difficult to obtain a phase balance with ordinary equipment within 2% over a band of audio frequencies. This means that carrier suppression is limited to a maximum of 40 db or so. However, when a relatively simple SSB transmitter is needed for spot frequency operation, a phasing unit will perform in a satisfactory manner. Where a high degree of performance in the SSB exciter is desired, the filter method and the phasing method may be combined. Through the use of the phasing method in the first balanced modulator those undesired sideband components lying within 1000 cycles of the carrier may be given a much higher degree of rejection than is attainable with the filter method alone, with any reasonable amount of complexity in the sideband filter. Then the sideband filter may be used in its normal way to attain very high attenuation of all undesired sideband components lying perhaps further than 500 cycles away from the carrier, and to restrict the sideband width on the desired side of the carrier to the specified frequency limit. Single Sideband Frequency Conversion Systems 17 -5 In many instances the band of sideband frequencies generated by a low level SSB transmitter must be heterodyned up to the desired carrier frequency. In receivers the circuits which perform this function are called converters or mixers. In sideband work they are usually termed mixers or modulators. Mixer Stages One circuit which can be used for this purpose employs a receiving -type mixer tube, such as the 6BE6. The output signal from the SSB generator is fed into the #1 grid and the conversion fre- www.americanradiohistory.com HANDBOOK Frequency Conversion 337 68E6 2000 KC. CONVERSION FREQUENCY (S.S ) 100 TUNE TO SELECT 2000 + 250 =2250 KC. ON 2000 -220 1750 KC. 250 KC. SSB SIGNAL (0.25V. ) SSB OUTPUT 12AU7 2.0 VOLT CONVERSION 0.2 VOLT SIGNAL INPUT SIGNAL Figure 27 PENTAGRID MIXER CIRCUIT FOR SSB FREQUENCY CONVERSION Figure 28 TWIN TRIODE MIXER CIRCUIT FOR quency into the #3 grid. This is the reverse of the usual grid connections, but it offers about 10 db improvement in distortion. The plate circuit is tuned to select the desired output frequency product. Actually, the output of the mixer tube contains all harmonics of the two input signals and all possible combinations of the sum and difference frequencies of all the harmonics. In order to avoid distortion of the SSB signal, it is fed to the mixer at a low level, such as 0.1 to 0.2 volts. The conversion frequency is fed in at a level about 20 db higher, or about 2 volts. By this means, harmonics of the incoming SSB signal generated in the mixer tube will be very low. Usually the desired output frequency is either the sum or the difference of the SSB generator carrier frequency and the conversion frequency. For example, using a SSB generator carrier frequency of 250 kc. and a conversion injection frequency of 2000 kc. as shown in figure 27, the output may be tuned to select either 2250 kc. or 1750 kc. Not only is it necessary to select the desired mixing product in the mixer output but also the undesired products must be highly attenuated to avoid having spurious output signals from the transmitter. In general, all spurious signals that appear within the assigned frequency channel should be at least 60 db below the desired signal, and those appearing outside of the assigned frequency channel at least 80 db below the signal level. When mixing 250 kc. with 2000 kc. as in the above example, the desired product is the 2250 kc. signal, but the 2000 kc. injection frequency will appear in the output about 20 db stronger than the desired signal. To reduce it to a level 80 db below the desired signal means that it must be attenuated 100 db. The principal advantage of using balanced modulator mixer stages is that the injection frequency theoretically does not appear in the output. In practice, when a considerable frequency range must be tuned by the balanced modulator and it is not practical to trim the SSB FREQUENCY CONVERSION push -pull circuits and the tubes into exact amplitude and phase balance, about 20 db of injection frequency cancellation is all that can be depended upon. With suitable trimming adjustments the cancellation can be made as high as 40 db, however, in fixed frequency circuit s. The Twin Triode Mixer The mixer circuit shown in figure 28 has about 10 db lower distortion than the conventional 6BE6 converter tube. It has a lower voltage gain of about unity and a lower output impedance which loads the first tuned circuit and reduces its selectivity. In some applications the lower gain is of no consequence but the lower distortion level is important enough to warrant its use in high performance equipment. The signal -to- distortion ratio of this mixer is of the order of 70 db compared to approximately 60 db for a 6BE6 mixer when the level of each of two tone signals is 0.5 volt. With stronger signals, the 6BE6 distortion increases very rapidly, whereas the 12AU7 distortion is much better comparatively. 6AS6's 001 SSB SIGNAL INPUT -BIAS CARRIER + 120 V. IN ATSMA. Figure 29 BALANCED MODULATOR CIRCUIT FOR SSB FREQUENCY CONVERSION 338 THE RADIO Sideband Transmission 1 7 6 Ell 3 ! .IéíNS IIIII1111EMY 9 . . . ... . . . . MURMUR ®UME=E__N 2:EN 8 7 6 BEM . 3 2 7 9 8 6 6 i 2 6 0o io 64 Figure 30 RESPONSE OF "N" NUMBER OF TUNED CIRCUITS, ASSUMING EACH CIRCUIT Q IS 50 www.americanradiohistory.com HANDBOOK Frequency Conversion In practical equipment where the injection frequency is variable and trimming adjustments and tube selection cannot be used, it may be easier and more economical to obtain this extra 20 db of attenuation by using an extra tuned circuit in the output than by using a balanced modulator circuit. A balanced modulator circuit of interest is shown in figure 29, providing a minimum of 20 db of carrier attenuation with no balancing adjustment. Selective Tuned Circuits The selectivity requirements of the tuned circuits following a mixer stage often become quite severe. For example, using an input signal at 250 kc. and a conversion injection frequency of 4000 kc. the desired output may be 4250 kc. Passing the 4250 kc. signal and the associated sidebands without attenuation and realizing 100 db of attenuation at 4000 kc. (which is only 250 kc. away) is a practical example. Adding the requirement that this selective circuit must tune from 2250 kc. to 4250 kc. further complicates the basic requirement. The best solution is to cascade a number of tuned circuits. Since a large number of such circuits may be required, the most practical solution is to use permeability tuning, with the circuits tracked together. An example of such circuitry is found in the Collins KWS -1 sideband transmitter. If an amplifier tube is placed between each tuned circuit, the overall response will be the sum of one stage multiplied by the number of stages (assuming identical tuned circuits). Figure 30 is a chart which may be used to determine the number of tuned circuits required for a certain degree of attenuation at some nearby frequency. The Q of the circuits is assumed to be 50, which is normally realized in small permeability tuned coils. The number of tuned circuits with a Q of 50 required for providing 100 db of attenuation at 4000 kc. while passing 4250 kc. may be found as fol- lows: of is 4250 -4000 =250 kc. fr is the resonant frequency, 4250 kc. and ff = 4250 - 0.059 The point on the chart where .059 intersects 100 db is between the curves for 6 and 7 tuned circuits, so 7 tuned circuits are required. Another point which must be considered in practice is the tuning and tracking error of the circuits. For example, if the circuits were 339 actually tuned to 4220 kc. instead of 4250 kc., the f'f would be 4220 or 0.0522. Checking the curves shows that 7 circuits would just barely provide 100 db of attenuation. This illustrates the need for very accurate tuning and tracking in circuits having high attenuation properties. Coupled Tuned When as many as 7 tuned circuits are required for proper attenuation, it is not necessary to have the gain that 6 isolating amplifier tubes would provide. Several vacuum tubes can be eliminated by using two or three coupled circuits between the amplifiers. With a coefficient of coupling between circuits 0.5 of critical coupling, the overall response is very nearly the same as isolated circuits. The gain through a pair of circuits having 0.5 coupling is only eight -tenths that of two critically coupled circuits, however. If critical coupling is used between two tuned circuits, the nose of the response curve is broadened and about 6 db is lost on the skirts of each pair of critically coupled circuits. In some cases it may be necessary to broaden the nose of the response curve to avoid adversely affecting the frequency response of the desired passband. Another tuned circuit may be required to make up for the loss of attenuation Circuits on the skirts of critically coupled circuits. Frequency Conversion Problems The example in the previous section shows the difficult selectivity problem encountered when strong undesired signals appear near the desired frequency. A high frequency SSB transmitter may be required to operate at any carrier frequency in the range of 1.75 Mc. to 30 Mc. The problem is to find a practical and economical means of heterodyning the generated SSB frequency to any carrier frequency in this range. There are many modulation products in the output of the mixer and a frequency scheme must be found that will not have undesired output of appreciable amplitude at or near the desired signal. When tuning across a frequency range some products may "cross over" the desired frequency. These undesired crossover frequencies should be at least 60 db below the desired signal to meet modern standards. The amplitude of the undesired products depends upon the particular characteristics of the mixer and the particular order of the product. In general, most products of the 7th order and higher will be at least 60 db down. Thus any cross- www.americanradiohistory.com THE RADIO Sideband Transmission 340 FROM SSB GENERATOR GAIN CONTROL PREAMPLIFIER y "SIGNAL TO DISTORTION (S /D) RATIO 3P 2V 2A-Q ANT. PLATE CIRCUIT R F RECTIFIER t 11 SP-4Q 4P-3R +TO STAGE R-F FROM P-A CONTROL BIAS II POWER AMPLIFIER I P Q 2Q7 3Q4j 431. DELAY BIAS VOLTAGE FROM POWER SUPPLY 5t41. Figure 31 SSB DISTORTION PRODUCTS, SHOWN UP TO NINTH ORDER Figure 32 BLOCK DIAGRAM OF AUTOMATIC LOAD CONTROL (A.L.C.) SYSTEM over frequency lower than the 7th must be avoided since there is no way of attenuating them if they appear within the desired pass band. The General Electric Ham News, volume 11 #6 of Nov. -Dec., 1956 covers the subject of spurious products and incorporates a "mix selector" chart that is useful in determining spurious products for various different mixing loaded, these spurious frequencies can extend far outside the original channel width and cause an unintelligible "splatter" type of interference in adjacent channels. This is usually of far more importance than the distortion of the original tones with regard to intelligibility or fidelity. To avoid interference in another channel, these distortion products should be down at least 40 db below adjacent channel signal. Using a two-tone test, the distortion is given as the ratio of the amplitude of one test tone to the amplitude of a third order product. This is called the signal -to- distortion ratio (S /D) and is usually given in decibels. The use of feedback r -f amplifiers make S/D ratios of greater than 40 db possible and practical. schemes. In general, for most applications when the intelligence bearing frequency is lower than the conversion frequency, it is desirable that the ratio of the two frequencies be between 5 to 1 and 10 to 1. This is a compromise between avoiding low order harmonics of this signal input appearing in the output, and minimizing the selectivity requirements of the circuits following the mixer stage. 17 -6 Distortion Products Due to Nonlinearity of R -F is Amplifiers When the SSB envelope of a voice signal distorted, a great many new frequencies are generated. These represent all of the possible combinations of the sum and difference frequencies of all harmonics of the original frequencies. For purposes of test and analysis, two equal amplitude tones are used as the SSB audio source. Since the SSB radio frequency amplifiers use tank circuits, all distortion products are filtered out except those which lie close to the desired frequencies. These are all odd order products; third order, fifth order, etc.. The third order products are 2p -q and 2q -p where p and q represent the two SSB r -f tone frequencies. The fifth order products are 3p -2q and 3q -2p. These and some higher order products are shown in figure 31. It should be noted that the frequency spacings are always equal to the difference frequency of the two original tones. Thus when a SSB amplifier is badly over- Two means may be used to keep the amplitude of these distortion products down to acceptable levels. One is to design the amplifier for excellent linearity over its amplitude or power range. The other is to employ a means of limiting the amplitude of the SSB envelope to the capabilities of the amplifier. An automatic load control ssytem (ALC) may be used to accomplish this result. It should be noted that the r.f wave shapes of the SSB signal are always sine waves because the tank circuits make them so. It is the change in gain with signal level in an amplifier that distorts the SSB envelope and generates unwanted distortion products. An ALC system may be used to limit the input signal to an amplifier to prevent a change in gain level caused by excessive input level. The ALC system is adjusted so the power amplifier is operating near its maximum power capability and at the same time is protected from being over -driven. In amplitude modulated systems it is common to use speech compressors and speech clipping systems to perform this function. These methods are not Automatic Load Control www.americanradiohistory.com HANDBOOK Distortion Products the signal is nearly up to the full power capability of the amplifier. At this signal level, the rectified output overcomes the delay bias and the gain of the preamplifier is reduced rapidly with increasing signal so that there is very little rise in output power above the threshold of gain control. When a signal peak arrives that would normally overload the power amplifier, it is desireable that the gain of the ALC amplifier be reduced in a few milliseconds to a value where overloading of the power amplifier is overcome. After the signal peak passes, the gain should return to the normal value in about one -tenth second. These attack and release times are commonly used for voice communications. For this type of work, a dynamic range of at least 10 db is desirable. Input peaks as high as 20 db above the threshold of compression should not cause loss of control although some increase in distortion in the upper range of compression can be tolerated because peaks in this range are infrequent. Another limitation is that the preceding SSB generator must be capable of passing signals above full power output by the amount of compression desired. Since the signal level through the SSB generator should be maintained within a limited range, it is unlikely that more than 12 db ALC action will be useful. If the input signal varies more than this, a speech compressor should be used to limit the range of the signal fed into the SSB generator. Figure 33 shows the effectiveness of the ALC in limiting the output signal to the capabilities of the power amplifier. An adjustment of the delay bias will place the threshold of compression at the desired power output. Figure 34 shows a simplified schematic of an ALC system. This ALC uses two variable gain am- DB SIGNAL LEVEL INPUT Figure 33 PERFORMANCE CURVE OF A.L.C. CIRCUIT equally useful in SSB. The reason for this is that the SSB envelope is different from the audio envelope and the SSB peaks do not necessarily correspond with the audio peaks as explained earlier in this chapter. For this reason a "compressor" of some sort located between the SSB generator and the power amplifier is most effective because it is controlled by SSB envelope peaks rather than audio peaks. Such a "SSB signal compressor" and the means of obtaining its control voltage comprises a satisfactory ALC system. The ALC Circuit A block diagram of an ALC circuit is shown in figure 32. The compressor or gain control part of this circuit uses one or two stages of remote cutoff tubes such as 6BA6, operating very similarly to the intermediate frequency stages of a receiver having automatic volume control. The grid bias voltage which controls the gain of the tubes is obtained from a voltage detector circuit connected to the power amplifier tube plate circuit. A large delay bias is used so that no gain reduction takes place until 6BA6 Figure 34 SIMPLIFIED SCHEMATIC OF AUTOMATIC LOAD CONTROL AMPLIFIER. OPERATING POINT OF ALC CIRCUIT MAY BE SET BY 6BA6 ee INPUT VARYING BLOCKING BIAS ON CATHODE OF 6X4 SIGNAL RECTIFIER SENS. ALC 341 =COMPRESSION INDICATOR ZERO ADJ. _ www.americanradiohistory.com 342 TH Sideband Transmission R E A D I O and through r -f filter capacitors. The 3.3K resistor and 0.1 µfd. capacitor across the rectifier output stabilizes the gain around the ALC loop to prevent "motor- boating." 17 -7 SSB JR. R -F and A -F Figure 35 MODULATOR CIRCUIT sources are applied to balanced modulator. in series plifier stages and the maximum overall gain is about 20 db. A meter is incorporated which is calibrated in db of compression. This is useful in adjusting the gain for the desired amount of load control. A capacity voltage divider is used to step down the r -f voltage at the plate of the amplifier tube to about 50 volts for the ALC rectifier. The output of the ALC rectifier passes through R -C networks to obtain the desired attack and release times Sideband Exciters Some of the most popular sideband exciters in use today are variations of the simple phasing circuit introduced in the November, 1950 issue of General Electric Ham Netes. Called the SSB, Jr., this simple exciter is the basis for many of the phasing transmitters now in use. Employing only three tubes, the SSB, Jr. is a classic example of sideband generation reduced to its simplest form. This phasing exciter employs audio and r -f phasing circuits to produce a SSB signal at one spot frequency. The circuit of one of the balanced modulator stages is shown in figure 35. The audio signal and r -f source are applied in series to two germanium diodes serving as balanced modulators The SSB, Jr. XTAL PHASE SHIFT NETWORK r T2 BLV FEO-WN/TE voF O. i 12AU7 AUDIO INPUT 000 RFC 250 ,, 0.5 MH f 101b 12ÁU7 I 2A17 TWIST 6AG7 -10.5V. C+,6- 6.3 V. G1, 2,3,4= INS GERMANIUM DIODE OR EQUIVALENT C2A,B,C,D =EACH SECTION 20LF, 450 V. ELECTROLYTIC C7= 2430 UUFD (.002 UFD MICA 5% WITH 170 -700 UUFD TRIMMER) LI, L2= 33 T. N21 E. WIRE CLOSEWOUND ON MILLEN N69046 IRON CORE ADJUSTABLE SLUG COIL FORM. LINK OF 6 CA =4600 UUFD. (.0043 UFD MICA 8.5% WITH 170-760 UUFD TRIMMER) TURNS OF HOOKUP WIRE WOUND ON OPEN END. C9= 1215 UUFD (.001 UFO MICA ±5% WITH 50 -360 UUFD TRIMMER) 5% WITH 5 -100 UUFD TRIMMER)L3 =16 T. N'19 E. WIRE SPACED TO FILL MILLEN M. 69046 C10 =607.5 UUFD (500 UUFD MICA FORM. TAP AT 6 TURNS. LINK OF TURN AT CENTER. COIL PARALLEL) AND 100UUFD MICA *10% (2S0UUFD 600V. C16= 35018UFD L4 =SAME AS L, EXCEPT NO LINK USED. 1% R7,RID= 133,300 OHMS, 1/2 WATT LS = 26 T. OF N19 E. WIRE. LINK ON END TO MATCH LOAD. 1% Re R9. 100,000 OHMS, 1/2 WATT (4 TURN LINK MATCHES 72 OHM LOAD) T1= STANCON A -53C TRANSFORMER. T2,T3= UTC R -364 TRANSFORMER. R- = MOUNTING ENO OF COILS S1= DPDT TOGGLE SWITCH f t 1 t t Figure 36 SCHEMATIC, SSB, JR. www.americanradiohistory.com HANDBOOK S.B. Exc iters having a push -pull output circuit tuned to the r -f "carrier" frequency. The modulator drives a linear amplifier directly at the output frequency. The complete circuit of the exciter is shown in figure 36. The first tube, a 12AU7, is a twin -triode serving as a speech amplifier and a crystal oscillator. The second tube is a I2AT7, acting as a twin channel audio amplifier following the phase-shift audio network. The linear amplifier stage is a 6AG7, capable of a peak power output of 5 watts. Sideband switching is accomplished by the reversal of audio polarity in one of the audio channels (switch SI), and provision is made for equalization of gain in the audio channels (R,z) . This adjustment is necessary in order to achieve normal sideband cancellation, which may be of the order of 35 db or better. Phase shift network adjustment may be achieved by adjusting potentiometer R5. Stable modulator balance is achieved by the balance potentiometers R,° and in conjunction with the germanium diodes. The SSB, Jr. is designed for spot frequency operation. Note that when changing frequency L,, L_, L.,, L, and L should be readjusted, since these circuits constitute the tuning adjustments of the rig. The principal effect of mistuning Ls, L., and L. will be lower output. The principal effect of mistuning L , however, will be degraded sideband suppression. Power requirements of the SSB, Jr. are 300 volts at 60 ma., and -10.5 volts at 1 ma. Under load the total plate current will rise to about 80 ma. at full level with a single tone input. With speech input, the total current will rise from the resting value of 60 ma. to about 70 ma., depending upon the voice waveform. R The "Ten -A" Exciter The Model 10 -A phasing exciter produced by Central Electronics, Inc. is an advanced version of the SSB, Jr. incorporating extra features such as VFO control, voice operation, and multi -band operation. A simplified schematic of the Model 10A is shown in figure 37. The 12AX7 two stage speech amplifier excites a transformer coupled 1/2 -12BH7 low impedance driver stage and a voice operated (VOX) relay system employing a 12AX7 and a 6AL5. A transformer coupled 12AT7 follows the audio phasing network, providing two audio channels having a 90- degree phase difference. A simple 90- degree r -f phase shift network in the plate circuit of the 9 Mc. crys- 343 tal oscillator stage works into the matched, balanced modulator consisting of four 1N48 diodes. The resulting 9 Mc. SSB signal may be converted to the desired operating frequency in a 6BA7 mixer stage. Eight volts of r-f from an external v -f -o injected on grid #1 of the 6BA7 is sufficient for good conversion efficiency and low distortion. The plate circuit of the 6BA7 is tuned to the sum or difference mixing frequency and the resulting signal is amplified in a 6AG7 linear amplifier stage. Two "tweet" traps are incorporated in the 6BA7 stage to reduce unwanted responses of the mixer which are apparent when the unit is operating in the 14 Mc. band. Band -changing is accomplished by changing coils L. and L° and the frequency of the external mixing signal. Maximum power output is of the order of 5 watts at any operating frequency. A Simple 80 Meter Phasing Exciter A SSB exciter employ - ing r -f and audio phasing circuits is shown in figure 38. Since the r -f phasing circuits are balanced only at one frequency of operation, the phasing exciter is necessarily a single frequency transmitter unless provisions are made to re- balance the phasing circuits every time a frequency shift is made. However for mobile operation, or spot frequency operation a relatively simple phasing exciter may be made to perform in a satisfactory manner. A 12AU7 is employed as a Pierce crystal oscillator, operating directly on the chosen SSB frequency in the 80 meter band. The second section of this tube is used as an isolation stage, with a tuned plate circuit, L. The output of the oscillator stage is link coupled to a 90° r -f phase -shift network wherein the audio signal from the audio phasing network is combined with the r -f signals. Carrier balance is accomplished by adjustment of the two 1000 ohm potentiometers in the r -f phase network. The output of the r -f phasing network is coupled through L_ to a single 6CL6 linear amplifier which delivers a 3 watt peak SSB signal on 80 meters. A cascade 12AT7 and a single 6C4 comprise the speech amplifier used to drive the audio phase shift network. A small inter -stage transformer is used to provide the necessary 180° audio phase shift required by the network. The output of the audio phasing network is coupled to a 12AU7 dual cathode follower which provides the necessary low impedance circuit to match the r -f phasing network. A double- www.americanradiohistory.com 344 Sideband Transmission THE RADIO Figure 37 SIMPLIFIED SCHEMATIC OF "TEN -A" EXCITER www.americanradiohistory.com HANDBOOK S.B. Exciters so* Boo K 12 AU) 111,8 345 6CL6 L3 L2 3 f100 K 3W PEAK SSB 150 Tuur C L1, L2,L38 24T0/22E.0w NOTE. UNLESS OTHERWISE SPECIFIED/ XR-30 PORN .001 (0.3. O/A.) CO/LS ARE T EACH. .001 ALL RESISTORS 0.3 WATT CARRIER EN-D4-1N71 INJECT. 1.0 70K 2511 SIDE BAN ALL CAPACITORS /N//P. ASTERISK AFTER CAPACITOR OR RESISTOR VALUE IND/CATES PRECIS /ON UNIT EAACT VALUE CRITICAL ONLY /N T/IAT /T SHOULD MATCH THE MAT/NG UNIT CLOSELY. D SELECTOR SWITCH 200 500K K 12AT7 STANCOR 6C 4 1 MIC. 2 12AÚ7 A53-C -5 14 7 M 10.2, r. SIMPLE 3 -WATT Figure 38 PHASING TYPE pole double -throw switch in the output circuit of the cathode follower permits sideband selection. A Filter -Type Exciter for 80 and 40 Meters A simple SSB filter - type exciter employing the Collins mechanical filter illustrates many of the basic principles of sideband generation. Such an exciter is shown in figure 39. The exciter is designed for operation in the 80 or 40 meter phone bands and delivers sufficient output to drive a class ABI tetrode such as the 2E26, 807, or 6146. A conversion crystal may be employed, or a separate conversion v -f -o can be used as indicated on the schematic illustration. The exciter employs five tubes, exclusive of power supply. They are: 6U8 low frequency oscillator and r -f phase inverter, 6BA6 i -f amplifier, 6BA7 high frequency mixer, 6AG7 linear amplifier, and 12AU7 speech amplifier and cathode follower. The heart of the exciter is the balanced modulator employing two 1N81 germanium diodes and the 455 kc., 3500 -cycle bandwidth mechanical filter. The input and output circuits of the filter are resonated to 455 kc. by means of small padding capacitors. A series -tuned Clapp oscillator covers the range of 452 kc. 457 kc. permitting the - +300 V. e 500 SSB EXCITER carrier frequency to be adjusted to the "20 decibel" points on the response curve of the filter, as shown in figure 40. Proper r -f signal balance to the diode modulator may be obtained by adjustment of the padding capacitor in the cathode circuit of the triode section of the 6U8 r -f tube. Carrier balance is set by means of a 50K potentiometer placed across the balanced modulator. One half of a 12AU7 serves as a speech amplifier delivering sufficient output from a high level crystal microphone to drive the second half of the tube as a low impedance cathode follower, which is coupled to the balanced modulator. The two 1N81 diodes act as an electronic switch, impressing a double sideband, suppressed- carrier signal upon the mechanical filter. By the proper choice of frequency of the beating oscillator, the unwanted sideband may be made to fall outside the pass band of the mechanical filter. Thus a single sideband suppressed- carrier signal appears at the output of the filter. The 455 kc. SSB signal is amplified by a 6BA6 pentode stage, and is then converted to a frequency in the 80 meter or 40 meter band by a 6BA7 mixer stage. Either a crystal or an external v -f -o may be used for the mixing signal. To reduce spurious signals, a double tuned www.americanradiohistory.com B.r w-EV , &MIN =WM p e WM ME OWE 'o I MOW I ,D 452 453 454 455 456 457 456 FREQUENCY (KC.) Figure 40 THE "TWENTY DB" CARRIER POINTS ON THE FILTER CURVE N Ç III ( 1,11 > , The beating oscillator p I IF O passbond. The carrier of the SSB signal is thus attenuated 20 db in addition to the OVN. inherent carrier attenuation of the balanced mixer. A total carrier attenuation of 50 db is achieved. Unwanted sideband rejection is of the some order. LL 3 In should be adjusted so that its frequency corresponds to the 20 db attenuation points of the mechanical filter " 6BE 6 R FC 500 RC. CARRIER INJECTION AUDIO SIGNAL 556 SIGNAL FROM F AMP. Figure 41 THE PRODUCT DETECTOR The above configuration resembles pentagrid converter circuit. ® ICYCLEOF - MWAV FORM ODULATIN ß. LYWER SIDEBAND UPPER SIDEBAND CARRIER FREQ. FREQUENCY SPECTRUM WITN COMPLEX MODULATING WAVE Figure 39 SCHEMATIC, FILTER -TYPE SSB EXCITER FOR 80 OR 40 METER OPERATION DOUBLE SIDE -BAND OUTPUT FROM BALANCED MODULATOR WITH SINE-WAVE MODULATION Figure 42 DOUBLE -SIDEBAND SUPPRESSED- CARRIER SIGNAL The envelope shown at B also is obtained on when two audio frequencies oscilloscope the of the some amplitude are fed to the input of a single -sideband transmitter. www.americanradiohistory.com S.B. transformer is placed between the mixer stage and the 6AG7 output stage. A maximum signal of 3 watts may be obtained from the 6AG7 linear amplifier. Selection of the upper or lower sideband is accomplished by tuning the 6U8 beating oscillator across the passband of the mechanical filter, as shown in figure 40. If the 80 meter conversion oscillator is placed on the low frequency side of the SSB signal, placing the 6U8 beating oscillator on the low frequency side of the passband of the mechanical filter will produce the upper sideband on 80 meters. When the beating oscillator is placed on the high frequency side of the passband of the mechanical filter the lower sideband will be generated on 80 meters. If the 80 meter conversion oscillator is placed on the high frequency side of the SSB signal, the sidebands will be reversed from the above. The variable oscillator should be set at approximately the 20 db suppression point of the passband of the mechanical filter for best operation, as shown in figure 40. If the oscillator is closer in frequency to the filter passband than this, carrier rejection will suffer. If the oscillator is moved farther away in frequency from the passband, the lower voice frequencies will be attenuated, and the SSB signal will sound high pitched and tinny. A little practice in setting the frequency of the beating oscillator while monitoring the 80 meter SSB signal in the station receiver will quickly acquaint the operator with the proper frequency setting of the beating oscillator control for transmission of either sideband. If desired, an amplitude modulated signal with full carrier and one sideband may be transmitted by placing the 6U8 low frequency oscillator just inside either edge of the passband of the filter (designated "AM point', figure 40) . After the 6U8 oscillator is operating over the proper frequency range it should be possible to tune the beating oscillator tuning capacitor across the passband of the mechanical filter and obtain a reading on the S -meter of a receiver tuned to the filter frequency and coupled to the input grid of the 6BA6 i -f amplifier tube. The two carrier balance controls of the 6U8 phase inverter section should be adjusted for a null reading of the S-meter when the oscillator is placed in the center of the filter passband. The 6BA6 stage is now checked for operation, and transformed T1 aligned to the carrier frequency. It may be necessary to unbalance temporarily potentio- Exciters 347 meter #2 of the 6U8 phase inverter in order to obtain a sufficiently strong signal for proper alignment of Ti. A conversion crystal is next plugged in the 6BA7 conversion oscillator circuit, and the operation of the oscillator is checked by monitoring the crystal frequency with a nearby receiver. The SSB "carrier" produced by the unbalance of potentiometer #2 should be heard at the proper sideband frequency in either the 80 meter or 40 meter band. The coupled circuit between the 6BA7 and the 6AG7 is resonated for maximum carrier voltage at the grid of the amplifier stage. Care should be taken that this circuit is tuned to the sideband frequency and not to the frequency of the conversion oscillator. Finally, the 6AG7 stage is tuned for maximum output. When these adjustments have been completed, the 455 kc. beating oscillator should be moved just out of the passband of the mechanical filter. The 80 meter "carrier" will disappear. If it does not, there is either energy leaking around the filter, or the amplifier stages are oscillating. Careful attention to shielding (and neutralization) should cure this difficulty. Audio excitation is now applied to the exciter, and the S-meter of the receiver should kick up with speech, but the audio output of the receiver should be unintelligible. As the frequency of the beating oscillator is adjusted so as to bring the oscillator frequency within the passband of the mechanical filter the modulation should become intelligible. A single sideband a.m. signal is now being generated. The BFO of the receiver should now be turned on, and the beating oscillator of the exciter moved out of the filter passband. When the receiver is correctly tuned, clean, crisp speech should be heard. The oscillator should be set at one of the "20 decibel" points of the filter curve, as shown in figure 40 and all adjustments trimmed for maximum carrier suppression. 17 -8 Reception of Single Sideband Signals Single-sideband signals may be received, after a certain degree of practice in the technique, in a quite adequate and satisfactory manner with a good communications receiver. However, the receiver must have quite good frequency stability both in the high- frequency oscillator and in the beat oscillator. For this reason, receivers which use a crystal -controlled first oscillator are likely to offer a www.americanradiohistory.com 348 TH Sideband Transmission greater degree of satisfaction than the more common type which uses a self -controlled oscillator. Beat oscillator stability in most receivers is usually quite adequate, but many receivers do not have a sufficient amplitude of beat oscillator injection to allow reception of strong SSB signals without distortion. In such receivers it is necessary either to increase the amount of beat- oscillator injection into the diode detector, or the manual gain control of the receiver must be turned down quite low. The tuning procedure for SSB signals is as follows: The SSB signals may first be located by tuning over the band with receiver set for the reception of c -w.; that is, with the manual gain at a moderate level and with the beat oscillator operating. By tuning in this manner SSB signals may be located when they are far below the amplitude of conventional AM signals on the frequency band. Then after a signal has been located, the beat oscillator should be turned off and the receiver put on a.v.c. Following this the receiver should be tuned for maximum swing of the S meter with modulation of the SSB signal. It will not be possible to understand the SSB signal at this time, but the receiver may be tuned for maximum deflection. Then the receiver is put back on manual gain control, the beat oscillator is turned on again, the manual gain is turned down until the background noise level is quite low, and the beat oscillator control is varied until the signal sounds natural. The procedure in the preceding paragraph may sound involved, but actually all the steps except the last one can be done in a moment. However, the last step is the one which will require some practice. In the first place, it is not known in advance whether the upper or lower sideband is being transmitted. So it will be best to start tuning the beat oscillator from one side of the pass band of the receiver to the other, rather than starting with the beat oscillator near the center of the pass band as is normal for c -w reception. With the beat oscillator on the wrong side of the sideband, the speech will sound inverted; that is to say that low- frequency modulation tones will have a high pitch and high- frequency modulation tones will have a low pitch and the speech will be quite unintelligible. With the beat oscillator on the correct side of the sideband but too far from the correct position, the speech will have some intelligibility but the voice will sound quite high pitched. Then as the correct setting for the beat oscilla- - E R A D I O tor is approached the voice will begin to sound natural but will have a background growl on each syllable. At the correct frequency for the beat oscillator the speech will clear completely and the voice will have a clean, crisp quality. It should also be mentioned that there is a narrow region of tuning of the beat oscillator a small distance on the wrong side of the side band where the voice will sound quite bassy and difficult to understand. With a little experience it will be possible to identify the sound associated with improper settings of the beat -oscillator control so that corrections in the setting of the control can be made. Note that the main tuning control of the receiver is not changed after the sideband once is tuned into the pass band of the receiver. All the fine tuning should be done with the beat oscillator control. Also, it is very important that the r -f gain control be turned to quite a low level during the tuning process. Then after the signal has been tuned properly the r -f gain may be increased for good signal level, or until the point is reached where best oscillator injection becomes insufficient and the signal begins to distort. Greatly simplified tuning, coupled with strong attenuation of undesired signals, can be obtained through the use of a single -sideband receiver or receiver adapter. The exalted carrier principle usually is employed in such receivers, with a phase -sensitive system sometimes included for locking the local oscillator to the frequency of the carrier of the incoming signal. In order for the locking system to operate, some carrier must be transmitted along with the SSB signal. Such receivers and adapters include a means for selecting the upper or lower sideband by the simple operation of a switch. For the reception of a single -sideband signal the switch obviously must be placed in the correct position. But for the reception of a conventional AM or phase- modulated signal, either sideband may be selected, allowing the sideband with the least interference to be used. Single -Sideband Receivers and Adopters An unusually satisfactory form of demodulator for SSB service is the product detector, shown in one form in figure 41. This circuit is preferred since it reduces intermodulation products and does not require a large local carrier voltage, as contrasted to the more common diode envelope detector. This product detector operates much in the same manner as The Product Detector www.americanradiohistory.com HANDBOOK S.S.B. Reception 4 -250A S4WS850 TURRET .001 ANT. 10 N V. 200 349 00= 605 MODULATORS 250 1500 IS KV. STANCOR A-782 O01 10Kv (!/SE PRIMARY AS SECONDARY 200 - h_ 4-250A ES 4000 V. Figure 43 HIGH -LEVEL DSB BALANCED MODULATOR a multi -grid mixer tube. The SSB signal is applied to the control grid of the tube and the locally generated carrier is impressed upon the other control grid. The desired audio output signal is recovered across the plate resistance of the demodulator tube. Since the cathode current of the tube is controlled by the simultaneous action of the two grids, the current will contain frequencies equal to the sum and difference between the sideband signal and the carrier. Other frequencies are suppressed by the low -pass r -f filter in the plate circuit of the stage, while the audio frequency is recovered from the i -f sideband signal. 17 -9 Double Sideband Transmission Many systems of intelligence transmission lie in the region between amplitude modulation on the one hand and single sideband suppressed- carrier transmission on the other hand. One system of interest to the amateur is the Synchronous Communications System, popularly known as "double sideband" (DSB -) transmission, wherein a suppressed -carrier double sideband signal is transmitted (figure 42) . Reception of such a signal is possible by utilizing a local oscillator phase -control system which derives carrier phase information from the sidebands alone and does not require the use of any pilot carrier. The DSB Transmitter 8 may be Synchronous Detection A DSB signal may be received with difficulty on a conventional receiver, a n d one of the two sidebands may easily be received on a single sideband receiver. For best reception, however, a phase- locked local oscillator and a synchronous detector should be employed. This operation may be performed either at the frequency of reception or at a convenient intermediate frequency. A block diaI -SYN- Low I-AUDIO FILTER AMPLIFIER I CHRONOUS DETECTOR PASS E- DS 8 SIGNAL LOCAL "OSCILLATOR FREQUENCY CONTROL AUDIO 0- DISCR PHASE IMIN. AUDIO AMPLIFIER V0 PHASE SHIFTER A balanced modulator of the type shown in employed to create a DSB signal. For higher operating levels, a pair of class -C type tetrode amplifier tubes may be screen modulated by a push -pull audio system figure and excited from a push -pull r -f source. The plates of such a modulator are connected in parallel to the tank circuit, as shown in figure 43. This DSB modulator is capable of 1 -kilowatt peak power output at a plate potential of 4000 volts. The circuit is self- neutralizing and the tune -up process is much the same as with any other class -C amplifier stage. As in the case of SSB, the DSB signal may also be generated at a low level and amplified in linear stages following the modulator. 0. -SYN- CHRONOUS DETECTOR Q -Low PASS FILTER Q -AUDIO AMPLIFIER Figure 44 BLOCK DIAGRAM OF DSB RECEIVING ADAPTER www.americanradiohistory.com 350 Sideband Transmission gram of a DSB synchronous receiver is shown in figure 44. The DSB signal is applied to two detectors having their local oscillator conversion voltages in phase quadrature to each other so that the audio contributions of the upper and lower sidebands reinforce one another. The in -phase oscillator voltage is adjusted to have the same phase as the suppressed carrier of the transmitted signal. The I- amplifier audio output, therefore, will contain the demodulated audio signal, while the Q- amplifier (supplied with quadrature oscillator voltage) will produce no output due to the quadrature null. Any frequency change of the local oscillator will produce some audio output in the Q-amplifier, while the I- amplifier is relatively unaffected. The Q- amplifier audio will have the same polarity as the I- channel audio for one direction of oscillator drift, and opposite polarity for oscillator drift in the opposite direction. The Q- amplifier signal level is proportional to the magnitude of the local oscillator phase angle error (the oscillator drift) for small errors. By combining the I- signal and the Q- signal in the audio phase discriminator a d -c control voltage is developed which automatically corrects for local oscillator phase errors. The reactance tube therefore locks the local oscillator to the correct phase. Phase control information is derived entirely from the sideband component of the signal and the carrier (if present) is not employed. Phase control ceases with no modulation of the signal and is reestablished with the reappearance of modulation. The Beam Deflection Modulator 17 -10 A recent development in the single side band field is the beam deflection tube (type 7360). This miniature tube employs a simple electron "gun" which generates, controls, and accelerates a beam of electrons directed toward identical plates. The total plate current is determined by the voltages applied to the control grid and screen grid of the "gun ". The division of plate current between the two plates is determined by the difference in voltage between two deflecting electrodes placed between the "gun" and the plates. R.f. voltage is used to modulate the control grid of the electron "gun" and the electron stream within the tube may be switched between the plates by means of an audio signal applied to the deflecting electrodes. The 7360 makes an excellent balanced modulator (figure 45) or product detector having high impedance input circuits, low distortion, and excellent carrier suppression. 7360 .00, I. SO Interference Rejection Interference falling within the passband of the receiver can be reduced by proper combination of the I- and Q- audio signals. Under phase lock conditions, the I- signal is composed of the audio signal plus the undesired interference, whereas the Q- signal contains only the interference component. Phase cancellation obtained by combining the two signals will reduce the interference while still adding the desired information contained in both side -bands. The degree of interference rejection is dependent upon the ratio of interference falling upon the two sidebands of the received signal and upon the basic design of the audio networks. A schematic and description of a complete DSB receiving adapter is shown in the June, 1957 issue of CQ magazine. R.F IN 470 K - PUSH-PULL AUDIO 1.265 V. Figure 45. BALANCED MODULATOR CIRCUIT USING 7360 BEAM DEFLECTION TUBE. www.americanradiohistory.com W V 2 CN - 0 ñ N o o a 0 0 0 N O ^ - 0 N V N O N 0 0 a N f1 - N O o -^ O N O N o 0 o o h 0 0 10 - N O N O N a Z X D 0 K Q O O O O O 0 a 0 O O 0 a o O 0 0 0 ^ ^ 0 0 0 ^ - 0 0 0 N N N N N a V O a a 0 O 0 0 0 0 o a O O O O O - O , O O O - O O N o- N - O .V O o o a O O - a 0 0 0 Ñ Ñ Ñ N O O O Ñ 0 0 0 0 a .- b 0 - o N O 0 0 0 0 0 0 0 .V N N Q i i i 05 IIIIIIIIIIIIIIIIIII;IIIII CO CC W a= ? Zl./ VI w CC O = U ln Z Z IU - 0 i_ W W 0 O Ç a o o? Ç a o o O- Ç a o 0 Ç O O O Ç a a O G O D F W a Z D O F 44 -14 - U 2 a2 ó a r O Ó á P O N N Q - Z X D Q NO.- r Ó Ó - - a a - 0 lai N NIO -IN - -IN eq N W 7 -0 "IV Ú 02 á F F 0 O a O O a N O F a- a N a *0 O N N NV O ti a O 1- 1- O O 0 O F F 0 0 O aN O .- p F N N 0 0 0 0 0 0 ö 0 a a N 0 p- F N a 0 O O O Ñ O - N N 0 0 a 0 0 N O F F 0 0 0 O F F 1N a 0 N O O F O o F O \ Jyp W a J 4 ó N N 0 0 0 0 0 0 0 0 0 0 0 0 N I CC w CL ZÚ crZ V 0 I N O 0ä r o 0 I a o N I O 80 N O o°° a N - No) o 0 I a 0 N I 0° N N N O- a I 0 01 I J ti2 vOW u 0 6. W '-IN 0I0 ed* www.americanradiohistory.com - ?óú F sail Z aN Jú ó? U til~ (Ei C4 NO 0° H OW R 0 o V W tw ¢_ 4 0N W Its' F O i Nu - úO 4? CHAPTER EIGHTEEN Transmitter Design The excellence of a transmitter is a function of the design, and is dependent upon the execution of the design and the proper choice of components. This chapter deals with the study of transmitter circuitry and of the basic components that go to make up this circuitry. Modern components are far from faultless. Resistors have inductance and distributed capacity. Capacitors have inductance and resistance, and inductors have resistance and distributed capacity. None of these residual attributes show up on circuit diagrams, yet they are as much responsible for the success or failure of the transmitter as are the necessary and vital bits of resistance, capacitance and inductance. Because of these unwanted attributes, the job of translating a circuit on paper into a working piece of equipment often becomes an impossible task to those individuals who disregard such important trivia. Rarely do circuit diagrams show such pitfalls as ground loops and residual inductive coupling between stages. Parasitic resonant circuits are rarely visible from a study of the schematic. Too many times radio equipment is rushed into service before it has been entirely checked. The immediate and only too apparent results of this enthusiasm are transmitter instability, difficulty of neutralization, r.f. wandering all over the equipment, and a general "touchiness" of adjustment. Hand in glove with these problems go the more serious ones of TVI, key -clicks, and parasitics. By paying attention to detail, with a good working knowledge of the limitations of the components, and with a basic conception of the actions of ground currents, the average amateur will be able to build equipment that will work "just like the book says." The twin problems of TVI and parasitics are an outgrowth of the major problem of overall circuit design. If close attention is paid to the cardinal points of circuitry design, the secondary problems of TVI and parasitics will in themselves be solved. 18-1 Resistors The resistance of a conductor is a function of the material, the form the material takes, the temperature of operation, and the frequency of the current passing through the resistance. In general, the variation in resistance due to temperature is directly proportional to the temperature change. With most wire -wound resistors, the resistance increases with temperature and returns to its original value when the temperature drops to normal. So- called composition or carbon resistors have less reliable temperature /resistance characteristics. They usually have a positive temperature coefficient, but the retrace curve as the resistor is cooled is often erratic, and in 352 www.americanradiohistory.com Resistors +3 353 +! +4 +4 R +3 A +2 Z Ñ +1 N_ UI w z W 2 i 0 0 -I 2 2 -3 U -S -30 -20 - 0 O 20 10 30 40 50 SO 70 !O SO b -20 100 DEGREES CENTIGRADE many cases the resistance does not return to heat cycle. It is for this reason that care must be taken when soldering composition resistors in circuits that require close control of the resistance value. Matched resistors used in phase- inverter service can be heated out of tolerance by the act of soldering them into the circuit. Long leads should be left on the resistors and a long nose pliers should grip the lead between the iron and the body of the resistor to act as a heat block. General temperature characteristics of typical carbon resistors are shown in figure 1. The behavior of an individual re- Figure Roc C- 10 20 30 40 50 !O 70 SO SO 100 HEAT CYCLE OF CONDITIONED COMPOSITION RESISTORS sistor will vary from these curves depending upon the manufacturer, the size and wattage of the resistor, etc. Inductance of Resistors Every resistor because of its physical size has in addition to its desired resistance, less desirable amounts of inductance and distributed capacitance. These quantities are illustrated in figure 2A, the general equivalent circuit of a resistor. This circuit represents the actual impedance network of a resistor at any frequency. At a certain specified frequency 2 a L -Ms--1001 0 1 HEAT CYCLE OF UNCONDITIONED COMPOSITION RESISTORS a O DEGREES CENTIGRADE Figure its original value after - 6-O 6 _"` ,--` `,__-_-` ``IZEIII-- ` _ __-=__-_"mg_ EQUIVALENT CIRCUIT OF A A RESISTOR 3 M. 2 0 S IO FREQUENCY (MC.) EQUIVALENT CIRCUIT OF A RESISTOR AT A PARTICULAR FREQUENCY Figure 3 FREQUENCY EFFECTS ON SAMPLE COMPOSITION RESISTORS IS 354 THE Transmitter Design . WEI! -"" i-1 R=55000n so s xo I =MINIM 20 i0 5 FREQUENCY (MC.) Figure 4 CURVES OF THE IMPEDANCE OF WIRE WOUND RESISTORS AT RADIO FREQUENCIES the impedance of the resistor may be thought of as a series reactance (X,) as shown in figure 2B. This reactance may be either inductive or capacitive depending upon whether the residual inductance or the distributed capacitance of the resistor is the dominating factor. As a rule, skin effect tends to increase the reactance with frequency, while the capacity between turns of a wire -wound resistor, or capacity between the granules of a composition resistor tends to cause the reactance and resistance to drop with frequency. The behavior of various types of composition resistors over a large frequency range is shown in figure 3. By proper component design, non -inductive resistors having a minimum of residual reactance characteristics may be constructed. Even these have reactive effects that cannot be ignored at high frequencies. Wirewound resistors act as low -Q inductors at radio frequencies. Figure 4 shows typical curves of the high frequency characteristics of cylindrical wirewound resistors. In addition to resistance variations wirewound resistors exhibit both capacitive and inductive reactance, depending upon the type of resistor and the operating frequency. In fact, such resistors perform in a fashion as low -Q r -f chokes below their parallel self -resonant frequency. 18-2 Capacitors The inherent residual characteristics of capacitors include series resistance, series inductance and shunt resistance, as shown in figure 5. The series resistance and inductance RsHUN' M; o--C L RADIO --- RSERIES Figure 5 EQUIVALENT CIRCUIT OF A CAPACITOR depend to a large extent upon the physical configuration of the capacitor and upon the material of which it is made. Of great interest to the amateur constructor is the series inductance of the capacitor. At a certain frequency the series inductive reactance of the capacitor and the capacitive reactance are equal and opposite, and the capacitor is in itself series resonant at this frequency. As the operating frequency of the circuit in which the capacitor is used is increased above the series resonant frequency, the effectiveness of the capacitor as a by-passing element deteriorates until the unit is about as effective as a block of wood. The usual forms of by -pass capacitors have dielectrics of paper, mica, or ceramic. For audio work, and low frequency r -f work up to perhaps 2 Mc. or so, the paper capacitors are satisfactory as their relatively high internal inductance has little effect upon the proper operation of the circuit. The actual amount of internal inductance will vary widely with the manufacturing process, and some types of paper capacitors have satisfactory characteristics up to a frequency of 5 Mc. or so. When considering the design of transmitting equipment, it must be remembered that while the transmitter is operating at some relatively low frequency of, say, 7 Mc., there will be harmonic currents flowing through the various by -pass capacitors of the order of 10 to 20 times the operating frequency. A capacitor that behaves properly at 7 Mc. however, may offer considerable impedance to the flow of these harmonic currents. For minimum harmonic generation and radiation, it is obviously of greatest importance to employ by-pass capacitors having the lowest possible internal By -Pass Capacitors inductance. Mica dielectric capacitors have much less internal inductance than do most paper condensers. Figure 6 lists self- resonant frequencies of various mica capacitors having various lead lengths. It can be seen from inspection of this table that most mica capacitors become self- resonant in the 12 -Mc. to 50 -Mc. region. The inductive reactance they would offer to harmonic currents of 100 Mc. or so www.americanradiohistory.com HANDBOOK CONDENSER .02 Capacitors LEAD LENGTHS RESONANT FREQ. LF MICA NONE 44.5 MC. .002 OF MICA NONE 23.5 MC. 10 MC. fT 55 MC. 24 MC 55 MC. ff 220 MC. NO MC. LF MICA .01 .000911F MICA .002 LF CERAMIC .001 500 LF CERAMIC 1111F BUTTON .001 LF CERAMIC .01 1JF CERAMIC ff NONE 14.5 MC. Figure 6 SELF -RESONANT FREQUENCIES OF VARIOUS CAPACITORS WITH RANDOM LEAD LENGTH would be of considerable magnitude. In certain instances it is possible to deliberately series resonate a mica capacitor to a certain frequency somewhat below its normal self -resonant frequency by trimming the leads to a critical length. This is sometimes done for maximum by- passing effect in the region of 40 Mc. to 60 Mc. The recently developed button -mica capacitors shown in figure 7 are especially designed to have extremely low internal inductance. Certain types of button -mica capacitors of small physical size have a self-resonant frequency in the region of 600 Mc. Ceramic dielectric capacitors in general have the lowest amount of series inductance per unit of capacitance of these three universally used types of by -pass capacitors. Typical resonant frequencies of various ceramic units are listed in figure 6. Ceramic capacitors are available in various voltage and capacity ratings and different physical configurations. Stand-off types such as shown in figure 7 are useful for by- passing socket and transformer terminals. Two of these capacitors may be mounted in close proximity on a chassis and connected together by an r -f choke to form a highly effective r-f filter. The inexpensive "clamshell" type of ceramic capacitor is recommended for general by- passing in r-f circuitry, as it is effective as a by -pass unit to well over 100 Mc. The large TV "doorknob" capacitors are useful as by -pass units for high voltage lines. These capacitors have a value of 500 micromicrofarads, and are available in voltage ratings up to 40,000 volts. The dielectric of these capacitors is usually titanium- dioxide. This material exhibits piezo -electric effects, and capacitors employing it for a dielectric will tend to "talk- back" when a -c voltages are applied across them. When these capaci- 355 tors are used as plate bypass units in a modulated transmitter they will cause acoustical noise. Otherwise they are excellent for general r -f work. A recent addition to the varied line of capacitors is the coaxial or " Hypass" type of capacitor. These capacitors exhibit superior by-passing qualities at frequencies up to 200 Mc. and the bulkhead type are especially effective when used to filter leads passing through partition walls between two stages. Variable Air Capacitors Even though air is the perfect dielectric, air capacitors exhibit losses because of the inherent resistance of the metallic parts that make up the capacitor. In addition, the leakage loss across the insulating supports may become of some consequence at high frequencies. Of greater concern is the inductance of the capacitor at high frequencies. Since the capacitor must be of finite size, it will have tie -rods and metallic braces and end plates, all of which contribute to the inductance of the unit. The actual amount of the inductance will depend upon the physical size of the capacitor and the methbd used to make contact to the stator and rotor plates. This inductance may be cut to a minimum value by using as small a capacitor as is practical, by using insulated tie rods to prevent the formation of closed inductive loops in the frame of the unit, and by making connections to the centers of the plate assemblies rather than to the ends as is com- .i y d Figure 7 TYPES OF CERAMIC AND MICA CAPACITORS SUITABLE FOR HIGH -FREQUENCY BYPASSING The Centralab 858S (1000 mad) is recoin. mended for screen and plate circuits of tetrode tubes. www.americanradiohistory.com 356 monly done. A large transmitting capacitor may have an inherent inductance as large as 0.1 microhenry, making the capacitor susceptible to parasitic resonances in the 50 Mc. to 150 Mc. range of frequencies. The question of optimum C/L ratiq and capacitor plate spacing is covered in Chapter Thirteen. For all -band operation of a high power stage, it is recommended that a capacitor just large enough for 40 -meter phone operation be chosen. (This will have sufficient capacitance for phone operation on all higher frequency bands.) Then use fixed padding capacitors for operation on 80 meters. Such padding capacitors are available in air, ceramic, and vacuum types. Specially designed variable capacitors are recommended for u -h -f work; ordinary capacitors often have "loops" in the metal frame which may resonate near the operating frequency. Variable vacuum capacitors because of their small physical size have less inherent inductance per unit of capacity than do variable air capacitors. Their losses are extremely low, and their dielectric strength is high. Because of increased production the cost of such units is now within the reach of the designer of amateur equipment, and their use is highly recommended in high power tank Variable Vacuum Capacitors circuits. 18 -3 THE Transmitter Design RADIO Tinned or stranded wire will show greater losses at these frequencies. Tank coil and tank capacitor leads should be of heavier wire than other r -f leads. The best type of flexible lead from the envelope of a tube to a terminal is thin copper strip, cut from thin sheet copper. Heavy, rigid leads to these terminals may crack the envelope glass when a tube heats or cools. Wires carrying only a.f. or d.c. should be chosen with the voltage and current in mind. Some of the low- filament- voltage transmitting tubes draw heavy current, and heavy wire must be used to avoid voltage drop. The voltage is low, and hence not much insulation is required. Filament and heater leads are usually twisted together. An initial check should be made on the filament voltage of all tubes of 25 watts or more plate dissipation rating. This voltage should be measured right at the tube sockets. If it is low, the filament transformer voltage should be raised. If this is impossible, heavier or parallel wires should be used for filament leads, cutting down their length if possible. Coaxial cable may be used for high voltage leads when it is desirable to shield them from r -f fields. RG -8 /U cable may be used at d -c potentials up to 8000 volts, and the lighter RG -17 /U may be used to potentials of 3000 volts. Spark -plug type high- tension wire may be used for unshielded leads, and will withstand 10,000 volts. If this cable is used, the high- voltage leads Wire and Inductors Any length of wire, no matter how short, has a certain value of inductance. This property is of great help in making coils and inductors, but may be of great hindrance when it is not taken into account in circuit design and construction. Connecting circuit elements (themselves having residual inductance) together with a conductor possessing additional inductance can often lead to puzzling difficulties. A piece of no. 10 copper wire ten inches long (a not uncommon length for a plate lead in a transmitter) can have a self- inductance of 0.15 microhenries. This inductance and that of the plate tuning capacitor together with the plate -to- ground capacity of the vacuum tube can form a resonant circuit which may lead to parasitic oscillations in the v -h -f regions. To keep the self- inductance at a minimum, all r -f carrying leads should be as short as possible and should be made out of as heavy material as possible. At the higher frequencies, solid enamelled copper wire is most efficient for r -f leads. may be cabled with filament and other low voltage leads. For high -voltage leads in low poc/ r exciters, where the plate voltage is not over 450 volts, ordinary radio hookup wire of good quality will serve the purpose. No r -f leads should be cabled; in fact it is better to use enamelled or bare copper wire for r -f leads and rely upon spacing for insulation. All r -f joints should be soldered, and the joint should be a good mechanical junction before solder is applied. The efficiency and Q of air coils commonly used in amateur equipment is a factor of the shape of the coil, the proximity of the coil to other objects (including the coil form) and the material of which the coil is made. Dielectric losses in so- called "air wound" coils are low and the Q of such coils runs in the neighborhood of 300 to 500 at medium frequencies. Unfortunately, most of the transmitting type plug -in coils on the market designed for link coupling have far too small a pick up link for proper operation at 7 Mc. and 3.5 Mc. The coefficient of coupling of these coils is about 0.5, and additional means must be employed to provide satisfactory coupling at these low frequencies. Additional inductance in series with the pick up link, the whole being reso- www.americanradiohistory.com HANDBOOK Rc Inductors L Rc 357 L "°-z' Rc L C ¡_ C DISTRIBUTED Figure ELECTRICAL EQUIVALENT OF R -F nated to the operating frequency will often permit satisfactory coupling. Coil Placement For best Q a coil should be in the form of a solenoid with length from one to two times the diameter. For minimum interstage coupling, coils should be made as small physically as is practicable. The coils should then be placed so that adjoining coils are oriented for minimum mutual coupling. To determine if this condition exists, apply the following test: the axis of one of the two coils must lie in the plane formed by the center turn of the other coil. If this condition is not met, there will be appreciable coupling unless the unshielded coils are very small in diameter or are spaced a considerable distance from each other. Insulation On frequencies above 7 Mc., cera- mic, polystyrene, or blycalex insulation is to be recommended. Cold flow must be considered when using polystyrene (Amphenol 912, etc.). Bakelite has low losses on the lower frequencies but should never be used in the field of high- frequency tank circuits. Lucite (or Plexiglas), which is available in rods, sheets, or tubing, is satisfactory for use at all radio frequencies where the r-J voltages are not especially high. It is very easy to work with ordinary tools and is not expensive. The loss factor depends to a considerable extent upon the amount and kind of plas- ticizer used. The most important thing to keep in mind regarding insulation is that the best insulation is air. If it is necessary to reinforce air -wound coils to keep turns from vibrating or touching, use strips of Lucite or polystyrene cemented in place with Amphenol 912 coil dope. This will result in lower losses than the commonly used celluloid ribs and Duco cement. 8 CHOKE AT VARIOUS FREQUENCIES At low frequencies, the distributed capacity has little effect and the electrical equivalent circuit of the r -f choke is as shown in figure 8A. As the operating frequency of the choke is raised the effect of the distributed capacity becomes more evident until at some particular frequency the distributed capacity resonates with the inductance of the choke and a parallel resonant circuit is formed. This point is shown in figure 8B. As the frequency of operation is further increased the overall reactance of the choke becomes capacitive, and finally a point of series resonance is reached (figure 8C.). This cycle repeats itself as the operating frequency is raised above the series resonant point, the impedance of the choke rapidly becoming lower on each successive cycle. A chart of this action is shown in figure 9. It can be seen that as the r -f choke approaches and leaves a condition of series resonance, the performance of the choke is seriously impaired. The condition of series resonance may easily be found by shorting the terminals of the r -f choke in question with a piece of wire and exploring the windings of the choke with a grid-dip oscillator. Most commercial transmitting type chokes have series resonances in the vicinity of 11 Mc. or 24 Mc. p.o :ill= , / 5 R -f chokes may be considered to be special inductances designed to have a high value of impedance over a large range of frequencies. A practical r -f choke has inductance, distributed capacitance, and resistance. Radio Frequency Chokes E 111111111V". 1111111/11116/11 /Iv IS 20 25 FREQUENCY (MC.) Figure 9 FREQUENCY- IMPEDANCE CHARACTERISTICS FOR TYPICAL PIE -WOUND R -F CHOKES www.americanradiohistory.com 358 LEAD NDUCTANCE Figure 10 GROUND LOOPS IN AMPLIFIER STAGES Using chassis return 8. Common ground point A. 18 -4 Grounds At frequencies of 30 Mc. and below, a chasmay be considered as a fixed ground refer- sis THE Transmitter Design ence, since its dimensions are only a fraction of a wavelength. As the frequency is increased above 30 Mc., the chassis must be considered as a conducting sheet on which there are points of maximum current and potential. However, for the lower amateur frequencies, an object may be assumed to be at ground potential when it is affixed to the chassis. In transmitter stages, two important current loops exist. One loop consists of the grid circuit and chassis return, and the other loop consists of the plate circuit and chassis return. These two loops are shown in figure 10A. It can be seen that the chassis forms a return for both the grid and plate circuits, and that ground currents flow in the chassis towards the cathode circuit of the stage. For some years the theory has been to separate these ground currents from the chassis by returning all ground leads to one point, usually the cathode of the tube for the stage in question. This is well and good if the ground leads are of minute length and do not introduce cross couplings between the leads. Such a technique is illustrated in figure 1013. wherein all stage components are grounded to the cathode pin of the stage socket. However, in transmitter RADIO construction the physical size of the components prevent such close grouping. It is necessary to spread the components of such a stage over a fairly large area. In this case it is best to ground items directly to the chassis at the nearest possible point, with short, direct grounding leads. The ground currents will flow from these points through the low inductance chassis to the cathode return of the stage. Components grounded on the top of the chassis have their ground currents flow through holes to the cathode circuit which is usually located on the bottom of the chassis, since such currents travel on the surface of the chassis. The usual "top to bottom" ground path is through the hole cut in the chassis for the tube socket. When the gain per stage is relatively low, or there are only a small number of stages on a chassis this universal grounding system is ideal. It is only in high gain stages (i -f strips) where the "gain per inch" is very high that circulating ground currents will cause operational instability. Intercoupling of It is important to prevent inGround Currents tercoupling of various different ground currents when the chassis is used as a common ground return. To keep this intercoupling at a minimum, the stage should be completely shielded. This will prevent external fields from generating spurious ground currents, and prevent the ground currents of the stage from upsetting the action of nearby stages. Since the ground currents travel on the surface of the metal, the stage should be enclosed in an electrically tight box. When this is done, all ground currents generated inside the box will remain in the box. The only possible means of escape for fundamental and harmonic currents are imperfections in this electrically tight box. Whenever we bring a wire lead into the box, make a ventilation hole, or bring a control shaft through the box we create an imperfection. It is important that the effect of these imperfections be reduced to a minimum. 18 -5 Holes, Leads and Shafts Large size holes for ventilation may be put in an electrically tight box provided they are properly screened. Perforated metal stock having many small, closely spaced holes is the best screening material. Copper wire screen may be used provided the screen wires are bonded together every few inches. As the wire corrodes, an insulating film prevents contact between the individual wires, and the attenuation of the screening suffers. The screening material should be carefully soldered to the www.americanradiohistory.com HANDBOOK 359 Shielding EXTERNAL FIELD TIN CAN BOTTOM WITH FLUTED EDGE PRESSED AGAINST PANEL HOLES FOR METER STUDS RUBBER GROMMET METER NUT COAXIAL SOCKET COAXIAL PLUG .001 CERAMIC RFC ,OOI PANEL PANEL METER METER LEAD CERAMIC PLUTEO EDGES TO MAKE 000D ELECTRICAL CONTACT WITH PANEL CENTER CONDUCTOR . RIGHT Figure 11A SIMPLE METER SHIELD -OREN- BOX HOLE box, or bolted with a spacing of not less than two inches between bolts. Mating surfaces of the box and the screening should be clean. A screened ventilation opening should be roughly three times the size of an equivalent unscreened opening, since the screening represents about a 70 per cent coverage of the area. Careful attention must be paid to equipment heating when an electrically tight box is used. Commercially available panels having half inch ventilating holes may be used as part of the box. These holes have much less attenuation than does screening, but will perform in a satisfactory manner in all but the areas of weakest TV reception. If it is desired to reduce leakage from these panels to a minimum, the back of the grille must be covered with screening tightly bonded to the panel. Doors may be placed in electrically tight boxes provided there is no r -f leakage around the seams of the door. Electronic weatherstripping or metal "finger stock" may be used to seal these doors. A long, narrow slot in a closed box has the tendency to act as a slot antenna and harmonic energy may pass more readily through such an opening than it would through a much larger circular hole. Variable capacitor shafts or switch shafts may act as antennas, picking up currents inside the box and re- radiating them outside of the box. It is necessary either to ground the shaft securely as it leaves the box, or else to make the shaft of some insulating material. A two or three inch panel meter requires a large leakage hole if it is mounted in the wall of an electrically tight box. To minimize leak age, the meter leads should be by- passed and shielded. The meter should be encased with a metal shield that makes contact to the box entirely around the meter. The connecting studs of the meter may project through the back of the metal shield. Such a shield may be made out of the end of a tin can of correct EXTERNAL FIELD INTERNAL GROUND J CURRENTS [[ LL (( OÑ EXTE0.1EORÓF BO1AlL CURRENTS WRONG Figure 11B of coaxial connectors on electrically tight box prevents escape of ground currents from interior of box. At the same time external fields are not conducted into the interior of the box. Use diameter, cut to fit the depth of the meter. This complete shield assembly is shown in figure 11A. Careful attention should be paid to leads entering and leaving the electrically tight box. Harmonic currents generated within the box can easily flow out of the box on power or control leads, or even on the outer shields of coaxially shielded wires. Figure 11B illustrates the correct method of bringing shielded cables into a box where it is desired to preserve the continuity of the shielding. Unshielded leads entering the box must be carefully filtered to prevent fundamental and harmonic energy from escaping down the lead. Combinations of r -f chokes and low inductance by -pass capacitors should be used in power leads. If the current in the lead is high, the chokes must be wound of large gauge wire. Composition resistors may be substituted for the r -f chokes in high impedance circuits. Bulkhead or feed - through type capacitors are preferable when passing a lead through a shield partition. A summary of lead leakage with various filter arrangements is shown in figure 12. Internal Leads Leads that connect two points within an electrically tight box www.americanradiohistory.com TEST FIELD STRENGTH IN UV NO. 12000 I THE Transmitter Design 360 2 10000 3 630 B SHIELDED OSCILLATOR ,o-SMALL HOLE IN SHIELD TO OSC. I CI WELDED HOOK-UP WIRE C2 RI 600 _2 150 S 6 70 7 140 R I z 600 110 10 c. C RFC RiC C4 RFC 25 TRACE 12 RFC I CI- x L--_1---- r-i J.c3 - MOOR CARBON RFC-OHMITE 2-50 R CI SO 75MLFCERAMIC FEED-THROUGH 4 RFC IEID _ D WZRE C2 -.005 DISC CERAMIC C3 - .01 SPRAGUE HI -PASS C4 - 005 CERAMIC FEED -THROUGH Figure 12 LEAD LEAKAGE WITH VARIOUS LEAD FILTERING SYSTEMS (COURTESY WIDBM) may pick up fundamental and harmonic currents if they are located in a strong field of flux. Any lead forming a closed loop with itself will pick up such currents, as shown in figure 13. This effect is enhanced if the lead happens to be self -resonant at the frequency at which the exciting energy is supplied. The solution for all of this is to by -pass all internal power leads and control leads at each end, and to shield these leads their entire length. All filament, bias, and meter leads should be so treated. This will make the job of filtering the leads as they leave the box much easier, since normally "cool" leads within the box will not have picked up spurious currents from nearby "hot" leads. was operated near this frequency marked instability was noted, and the filaments of the 810 tubes increased in brilliance when plate voltage was applied to the amplifier, indicating the presence of r.f. in the filament circuit. Changing the filament by -pass capacitors to .01-pfd. lowered the filament resonance frequency to 2.2 Mc. and cured this effect. A ceramic capacitor of .01 -pfd. used as a filament by -pass capacitor on each filament leg seems to be satisfactory from both a resonant and a TVI point of view. Filament by -pass capacitors smaller in value than .01-pfd. should be used with caution. Various parasitic resonances are also found in plate and grid tank circuits. Push -pull tank circuits are prone to double resonances, as shown in figure 14. The parasitic resonance circuit is usually several megacycles higher than the actual resonant frequency of the full tank circuit. The cure for such a double resonance is the inclusion of an r -f choke in the center tap lead to the split coil. From a point of view of electrical properties, aluminum is a poor chassis material. It is difficult to make a soldered joint to it, and all grounds must rely upon a pressure joint. These presChassis Material Figure 13 SHIELDED SHIELDED COMPARTMENT COMPARTMENT RADIATION FIELD \ \. HOLE ICKU' LOOP RE-RADIATED FIELD 1 BY-PASS CAPACITOR BY -PASS CAPACITOR ILLUSTRATION OF HOW A SUPPOSEDLY GROUNDED POWER LEAD CAN COUPLE ENERGY FROM ONE COMPARTMENT TO ANOTHER OMPARTMENT Parasitic Resonances Filament leads within vacuum tubes may resonate with the filament by -pass capacitors at some particular frequency and cause instability in an amplifier stage. Large tubes of the 810 and 250TH type are prone to this spurious effect. In particular, a push -pull 810 amplifier using .001 -µfd. filament by -pass capacitors had a filament resonant loop that fell in the 7 -Mc. amateur band. When the amplifier WRONG RADIATION LOOP LECTRICALLY -TIGHT 18 -6 RADIO \\ RADIATION FIELD R IGHT ELECTRICALLY -TIGHT COMPARTMENT BULKHEAD TYPE fBY -PASS CAPACITOR ILLUSTRATION OF LEAD ISOLATION BY PROPER USE OF BULKHEAD BYPASS CAPACITOR www.americanradiohistory.com Parasitic Oscillations HANDBOOK RIGHT WRONG Figure 361 final amplifier stage that might be very severe if the plate voltage were left on and the excitation were keyed. In some cases, an all -wave receiver will prove helpful in locating v -h -f spurious oscillations, but it may be necessary to check from several hundred megacycles downward in frequency to the operating range. A normal harmonic is weaker than the fundamental but of good tone; a strong harmonic or a rough note at any frequency generally indicates a para- sitic. 14 DOUBLE RESONANCE EFFECTS IN PUSH PULL TANK CIRCUIT MAY BE ELIMINATED BY THE INSERTION OF ANY R -F CHOKE IN THE COIL CENTER TAP LEAD In general, the cure for parasitic oscillation is two-fold: The oscillatory circuit is damped until sustained oscillation is impossible, or it is detuned until oscillation ceases. An examination of the various types of parasitic oscillations and of the parasitic oscillatory circuits will prove handy in applying the correct cure. later date because of high resistivity caused by the formation of oxides from eletrolytic action in the joint. However, the ease of working sure joints are prone to give trouble at a and forming the aluminum material far outweighs the electrical shortcomings, and aluminum chassis and shielding may be used with good results provided care is taken in making all grounding connections. Cadmium and zinc plated chassis are preferable from a corrosion standpoint, but are much more difficult to handle in the home workshop. 18 -7 Parasitic Oscillation in R -F Amplifiers Parasitics (as distinguished from sell- oscillation on the normal tuned frequency of the amplifier) are undesirable oscillations either of very high or very low frequencies which may occur in radio -frequency amplifiers. They may cause spurious signals (which are often rough in tone) other than normal harmonics, hash on each side of a modulated carrier, key clicks, voltage breakdown or flashover, instability or inefficiency, and shortened life or failure of the tubes. They may be damped and stop by themselves after keying or modulation peaks, or they may be undamped and build up during ordinary unmodulated transmission, continuing if the excitation is removed. They may result from series or parallel resonant circuits of all types. Due to neutralizing lead length and the nature of most parasitic circuits, the amplifier usually is not neutralized for the parasitic frequency. Sometimes the fact that the plate supply is keyed will obscure parasitic oscillations in a Low Frequency One type of unwanted oscillation often occurs in shunt -fed circuits in which the grid and plate chokes resonate, coupled through the tube's interelectrode capacitance. This also can happen with series feed. This oscillation is generally at a much lower frequency than the operating frequency and will cause additional carriers to appear, spaced Parasitic Oscillations from perhaps twenty to a few hundred kilocycles on either side of the main wave. Such a circuit is illustrated in figure 15. In this case, RFC, and RFC2 form the grid and plate inductances of the parasitic oscillator. The neutralizing capacitor, no longer providing out -ofphase feedback to the grid circuit actually enhances the low frequency oscillation. Because of the low Q of the r -f chokes, they will usually run warm when this type of parasitic oscillation is present and may actually char and burn up. A neon bulb held near the oscillatory circuit will glow a bright yellow, the color appearing near the glass of the neon bulb and not between the electrodes. One cure for this type of oscillation is to change the type of choke in either the plate or the grid circuit. This is a marginal cure, because the amplifier may again break into the same type of oscillation when the plate voltage is raised slightly. The best cure is to remove the grid r -f choke entirely and replace it with a wirewound resistor of sufficient watt- age to carry the amplifier grid current. If the inclusion of such a resistor upsets the operating bias of the stage, an r -f choke may be used, with a 100 -ohm 2 -watt carbon resistor in series with the choke to lower the operating Q of the choke. If this expedient does not eliminate the condition, and the stage under investigation uses a beam -tetrode tube, negative resistance can exist in the screen circuit www.americanradiohistory.com THE Transmitter Design 362 RADIO 2 RFC2 RFC R F. RFC! RFCz GRID vLATE TANIÇ CURE PARASITIC CIRCUIT FOR LOW FREQ. OSCILLATION CIRCUIT Figure THE CAUSE AND CURE 15 OF LOW of such tubes. Try larger and smaller screen by -pass capacitors to determine whether or not they have any effect. If the condition is corning from the screen circuit an audio choke with a resistor across it in series with the screen feed lead will often eliminate the trouble. Low -frequency parasitic oscillations can often take place in the audio system of an AM transmitter, and their presence will not be known until the transmitter is checked on a receiver. It is easy to determine whether or not the oscillations are coming from the modulator simply by switching off the modulator tubes. If the oscillations are coming from the modulator, the stage in which they are being generated can be determined by removing tubes successively, starting with the first speech amplification stage, until the oscillation stops. When the stage has been found, remedial steps can be taken on that stage. If the stage causing the oscillation is a lowlevel speech stage it is possible that the trouble is coming from r -f or power- supply feedback, or it may be coming about as a result of inductive coupling between two transformers. If the oscillation is taking place in a high -level audio stage, it is possible that inductive or capacitive coupling is taking place back to one of the low -level speech stages. It is also possible, in certain cases, that parasitic push -pull oscillation can take place in a Class B or Class AB modulator as a result of the grid -to-plate capacitance within the tubes and in the stage wiring. This condition is more likely to occur if capacitors have been placed across the secondary of the driver transformer and across the primary of the modulation transformer to act in the reduction of the amplitude of the higher audio frequencies. Relocation of wiring or actual neutralization of the audio stage in the manner used for r -f stages may be required. FREQUENCY PARASITICS It may be said in general that the presence low- frequency parasitics indicates that somewhere in the oscillating circuit there is an impedance which is high at a frequency in the upper audio or low r -f range. This impedance may include one or more r -f chokes of the conventional variety, power supply chokes, modulation components, or the high impedance may be presented simply by an RC circuit such as might be found in the screen -feed circuit of a beam -tetrode amplifier stage. of 18 -8 Elimination of V -H -F Parasitic Oscillations V -h -f parasitic oscillations are often difficult to locate and difficult to eliminate since their frequency often is only moderately above the desired frequency of operation. But it may be said that v -h -f parasitics always may be eliminated if the operating frequency is appreciably below the upper frequency limit for the tubes used in the stage. However, the elimination of a persistent parasitic oscillation on a frequency only moderately higher than the desired operating frequency will involve a sacrifice in either the power output or the power sensitivity of the stage, or in both. Beam- tetrode stages, particularly those using 807 type tubes, will almost invariably have one or more v -h -f parasitic oscillations unless adequate precautions have been taken in advance. Many of the units described in the constructional section of this edition had parasitic oscillations when first constructed. But these oscillations were eliminated in each case; hence, the expedients used in these equipments should be studied. V-h -f parasitics may be readily identified, as they cause a www.americanradiohistory.com Parasitic Oscillations HANDBOOK 363 neon lamp to have a purple glow close to the electrodes when it is excited by the parasitic energy. Parasitic Oscillations with Triodes Triode stages are less subject to parasitic os- cillations primarily because of the much lower power sensitivity of such tubes as compared to beam tetrodes. But such oscillations can and do take place. Usually, however, it is not necessary to incorporate losser resistors as normally is the case with beam tetrodes, unless the triodes are operated quite near to their upper frequency limit, or the tubes are characterized by a relatively high transconductance. Triode v -h -f parasitic oscillations normally may be eliminated by adjustment of the lengths and effective inductance of the leads to the elements of the tubes. In the case of triodes, v -h -f parasitic oscillations often come about as a result of inductance in the neutralizing leads. This is particularly true in the case of push -pull amplifiers. The cure for this effect will usually be found in reducing the length of the neutralizing leads and increasing their diameter. Both the reduction in length and increase in diameter will reduce the inductance of the leads and tend to raise the parasitic oscillation frequency until it is out of the range at which the tubes will oscillate. The use of straightforward cir- cuit design with short leads will assist in forestalling this trouble at the outset. Butterfly -type tank capacitors with the neutralizing capacitors built into the unit (such as the B &W type) are effective in this regard. V-h -f parasitic oscillations may take place as a result of inadequate by-passing or long by -pass leads in the filament, grid- return and plate -return circuits. Such oscillations also can take place when long leads exist between the grids and the grid tuning capacitor or between the plates and the plate tuning capacitor. The grid and plate leads should be kept short, but the leads from the tuning capacitors to the tank coils can be of any reasonable length insofar as parasitic oscillations are concerned. In an amplifier where oscillations have been traced to the grid or plate leads, their elimination can often be effected by making the grid leads much longer than the plate leads or vice versa. Sometimes parasitic oscillations can be eliminated by using iron or nichrome wire for the grid or plate leads, or for the neutralizing leads. But in any event it will always be found best to make the neutralizing leads as short and of as heavy conductor as is practicable. In cases where it has been found that increased length in the grid leads for an amplifier is required, this increased length can often be wound into the form of a small coil and still Figure 16 GRID PARASITIC SUPPRESSORS IN PUSH PULL TRIODE STAGE obtain the desired effect. Winding these small coils of iron or nichrome wire may sometimes be of assistance. To increase losses at the parasitic frequency, the parasitic coils may be wound on 100 -ohm resistors. These "lossy" suppressors should be placed in the grid leads of the tubes close to the grid connection, as shown in fig2 -watt ure 16. Parasitics with Where beam -tetrode tubes are used in the stage which has been found to be generating the parasitic oscillation, all the foregoing suggestions apply in general. However, there Beam Tetrodes are certain additional considerations involved in elimination of parasitics from beam -tetrode amplifier stages. These considerations involve the facts that a beam -tetrode amplifier stage has greater power sensitivity than an equivalent triode amplifier, such a stage has a certain amount of screen -lead inductance which may give rise to trouble, and such stages have a small amount of feedback capacitance. Beam - tetrode stages often will require the inclusion of a neutralizing circuit to eliminate oscillation on the operating frequency. However, oscillation on the operating frequency normally is not called a parasitic oscillation, and different measures are required to eliminate the condition. Basically, parasitic oscillations in beam tetrode amplifier stages fall into two classes: cathode -grid- screen oscillations, and cathode screen -plate oscillations. Both these types of oscillation can be eliminated through the use of a parasitic suppressor in the lead between the screen terminal of the tube and the screen by -pass suppressor, as shown in figure 17. Such a suppressor has negligible effect on the by- passing effect of the screen at the operating frequency. The method of connecting this www.americanradiohistory.com 364 THE Transmitter Design PC=ST *18E. RADIO ON 52/1, 2W CAR- BON RESISTOR RFC=ONM/TE Z50 OR EQUIVALENT Figure 17 PARASITIC SUPPRESSION CIRCUIT FOR TETRODE TUBES SCREEN suppressor to tubes having dual screen leads is shown in figure 18. At the higher frequencies at which parasitics occur, the screen is no longer at ground potential. It is therefore necessary to include an r -f choke by-pass condenser filter in the screen lead after the parasitic suppressor. The screen lead, in addition, should be shielded for best results. During parasitic oscillations, considerable r -f voltage appears on the screen of a tetrode tube, and the screen by -pass condenser can easily be damaged. It is best, therefore, to employ screen by -pass condensers whose d -c working voltage is equal to twice the maximum applied screen voltage. The grid- screen oscillations may occasionally be eliminated through the use of a parasitic suppressor in series with the grid lead of the tube. The screen plate oscillations may also be eliminated by inclusion of a parasitic suppressor in series with the plate lead of the tube. A suitable grid suppressor may be made of a 22 -ohm 2 -watt Ohmite or Allen- Bradley resistor wound with 8 turns of no. 18 enameled wire. A plate circuit suppressor is more of a problem, since it must dissipate a quantity of power that is dependent upon just how close the parasitic frequency is to the operating frequency of the tube. If the two frequencies are close, the suppressor will absorb some of the fundamental plate circuit power. For kilowatt stages operating no higher than 30 Mc. a satisfactory plate circuit suppressor may be made of five 570 -ohm 2 -watt carbon resistors in parallel, shunted by 5 turns of no. 16 enameled wire, % inch diameter and % inch long (figure 19A and B). The parasitic suppressor for the plate circuit of a small tube such as the 5763, 2E26, 807, 6146 or similar type normally may consist of a 47 -ohm carbon resistor of 2 -watt size with 6 turns of no. 18 enameled wire wound around the resistor. However, for operation above 30 Mc., special tailoring of the value lj ""``""ftmeYrnrw. Figure 18 APPLICATION OF SCREEN PARASITIC SUPPRESSION CIRCUIT PHOTO OF OF FIGURE 17 of the resistor and the size of the coil wound around it will be required in order to attain satisfactory parasitic suppression without excessive power loss in the parasitic suppressor. Isolation between the grid and plate circuits of a tetrode tube is not perfect. For maximum stability, it is recommended that the tetrode stage be neutralized. Neutralization is absolutely necessary unless the grid and plate circuits Tetrode Screening of the tetrode stage are each completely isolated from each other in electrically tight boxes. Even when this is done, the stage will show signs of regeneration when the plate and grid tank circuits are tuned to the same frequency. Neutralization will eliminate this regeneration. Any of the neutralization circuits described in the chapter Generation of R -F Energy may be used. 18 -9 Checking for -Parasitic Oscillations It is an unusual transmitter which harbors no parasitic oscillations when first constructed www.americanradiohistory.com Parasitic Oscillations HANDBOOK 365 PC PC FOR PC =! T PC R/eE. ON 22d.2W. COM- 507, ETC. =ArI /IE.ON47/1,2W POSITION RESISTOR COMPOSITION RESISTOR FOR 4 PC = S -250A, ETC. -570!2, 2W COMPOSITION RESISTORS IN PARALLEL WITH ST. 0 /eE. //4. O Figure 19 PLATE AND GRID PARASITIC SUPPRESSION IN TETRODE TUBES and tested. Therefore it is always wise to follow a definite procedure in checking a new transmitter for parasitic oscillations. Parasitic oscillations of all types are most easily found when the stage in question is running by itself, with full plate (and screen) voltage, sufficient protective bias to limit the plate current to a safe value, and no excitation. One stage should be tested at a time, and the complete transmitter should never be put on the air until all stages have been thoroughly checked for parasitics. To protect tetrode tubes during tests for parasitics, the screen voltage should be applied through a series resistor which will limit the screen current to a safe value in case the plate voltage of the tetrode is suddenly removed when the screen supply is on. The correct procedure for parasitic testing is as follows (figure 20): 1. The stage in question should be coupled to a dumpy load, and tuned up in correct oper- ating shape. Sufficient protective bias should be applied to the tube at all times. For protection of the stage under test, a lamp bulb should be added in series with one leg of the primary circuit of the high voltage power supply. As the plate supply load increases during a period of parasitic oscillation, the voltage drop across the lamp increases, and the effective plate voltage drops. Bulbs of various size may be tried to adjust the voltage under testing conditions to the correct amount. If a Variac or Powerstat is at hand, it may be used in place of the bulbs for smoother voltage control. Don't test for parasitics unless some type of voltage control is used on the high voltage supply! When a stage breaks into parasitic oscillations, the plate current increases violently, and some protection to the tube under test must be used. 2. The r -f excitation to the tube should now be removed. When this is done, the grid, screen and plate currents of the tube should drop to zero. Grid and plate tuning condensers should be tuned to minimum capacity. No change in resting grid, screen or plate current should be observed. If a parasitic is present, grid current will flow, and there will be an abrupt increase in plate current. The size of the lamp bulb in series with the high voltage supply may be varied until the stage can oscillate contin- uously, without exceeding the rated plate dissipation of the tube. 3. The frequency of the parasitic may now be determined by means of an absorption wave meter, or a neon bulb. Low frequency oscillations will cause a neon bulb to glow yellow. High frequency oscillations will cause the bulb to have a soft, violet glow. Once the frequency of oscillation is determined, the cures suggested in this chapter may be applied to the stage. 4. When the stage can pass the above test with no signs of parasitics, the bias supply of the tube in question should be decreased until the tube is dissipating its full plate rating when full plate voltage is applied, with no r-f TO BE TESTED FOR EXCITER CONTROL SWITCH nl AMPLIFIER STAGE EXCITER PARASITICS DUMMY LOAD ° BIAS SUPPLY HIGH VOLTAGE POWER SUPPLY VARIAC OR LIGHT BULBS Figure 20 SUGGESTED TEST SETUP FOR PARASITIC TESTS www.americanradiohistory.com 4 A.C. SUPPLY 366 Transmitter Design excitation. Excitation may now be applied and the stage loaded to full input into a dummy load. The signal should now be monitored in a nearby receiver which has the antenna terminals grounded or otherwise shorted out. A series of rapid dots should be sent, and the frequency spectrum for several megacycles each side of the carrier frequency carefully searched. If any vestige of parasitic is left, it will show up as an occasional "pop" on a keyed dot. This "pop" may be enhanced by a slight detuning of either the grid or plate circuit. 5. If such a parasitic shows up, it means that the stage is still not stable, and further measures must be applied to the circuit. Parasitic suppressors may be needed in both screen and grid leads of a tetrode, or perhaps in both grid and neutralizing leads of a triode stage. As a last resort, a 10,000 -ohm 25 -watt wire wound resistor may be shunted across the grid coil, or grid tuning condenser of a high powered stage. This strategy removed a keying pop that showed up in a commercial transmitter, operating at a plate voltage of 5000. Test for Parasitic Tendency in Tetrodo Amplifiers is common experience to develop an engineer ing model of a new It equipment that is apparently free of parasitics and then find troublesome oscillations showing up in production units. The reason for this is that the equipment has a parasitic tendency that remains below the verge of oscillation until some change in a component, tube gain, or operating condition raises the gain of the parasitic circuit enough to start oscillation. In most high frequency transmitters there are a great many resonances in the tank circuits at frequencies other than the desired SIGNAL GENERATOR 100CC -20O MC Figure 21 PARASITIC GAIN MEASUREMENT Grid -dip oscillator and vacuum tube voltmeter may be used to measure para- sitic stage gain over region. IOOkc -200mc operating frequency. Most of these parasitic resonant circuits are not coupled to the tube and have no significant tendency to oscillate. A few, however, are coupled to the tube in some form of oscillatory circuit. If the regeneration is great enough, oscillation at the parasitic frequency results. Those spurious circuits existing just below oscillation must be found and suppressed to a safe level. One test method is to feed a signal from a grid -dip oscillator into the grid of a stage and measure the resulting signal level in the plate circuit of the stage, as shown in figure 21. The test is made with all operating voltages applied to the tubes. Class C stages should have bias reduced so a reasonable amount of static plate current flows. The grid -dip oscillator is tuned over the range of 100 kc to 200 mc. and the relative level of the r -f voltmeter is watched and the frequencies at which voltage peaks occur are noted. Each significant peak in voltage gain in the stage must be investigated. Circuit changes or suppression must then be added to reduce all peaks by 10 db or more in amplitude. www.americanradiohistory.com CHAPTER NINETEEN Television and Broadcast Interference The problem of interference to television reception is best approached by the philosophy discussed in Chapter Eighteen. By correct design procedure, spurious harmonic generation in low frequency transmitters may be held to a minimum. The remaining problem is twofold: to make sure that the residual harmonics generated by the transmitter are not radiated, and to make sure that the fundamental signal of the transmitter does not overload the television receiver by reason of the proximity of one to the other. In an area of high TV- signal field intensity the TVI problem is capable of complete solution with routine measures both at the amateur transmitter and at the affected receivers. But in fringe areas of low TV- signal field strength the complete elimination of TVI is a difficult and challenging problem. The fundamentals illustrated in Chapter Fifteen must be closely followed, and additional antenna filtering of the transmitter is required. 19 -1 Even if the amateur transmitter were perfect and had no harmonic radiation or spurious emissions whatever, it still would be likely to cause overloading of TV sets whose antennas were within a few hundred feet of the transmitting antenna. This type of overloading is essentially the same as the common type of BCI encountered when operating a medium power or high -power amateur transmitter within a few hundred feet of the normal type of BCL receiver. The field intensity in the immediate vicinity of the transmitting antenna is sufficiently high that the amateur signal will get into the BC or TV set either through overloading of the front end, or through the i -f, video, or audio system. A characteristic TV Set Overloading of this type of interference is that it always will be eliminated when the transmitter temporarily is operated into a dummy antenna. Another characteristic of this type of overloading is that its effects will be substantially continuous over the entire frequency coverage of the BC or TV receiver. Channels 2 through 13 will be affected in approximately the same manner. With the overloading type of interference the problem is simply to keep the fundamental of the transmitter out of the affected receiver. Other types of interference may or may not show up when the fundamental is taken out of the TV set (they probably will appear), but at least the fundamental must be eliminated first. The elimination of the transmitter fundamental from the TV set is normally the only operation performed on or in the vicinity of the TV receiver. After the fundamental has been elimi- Types of Television Interference There are three main types of TVI which may be caused singly or in combination by the emissions from an amateur transmitter. These types of interference are: (1) Overloading of the TV set by the trans(2) (3) mitter fundamental Impairment of the picture by spurious emissions Impairment of the picture by the radiation of harmonics . 367 www.americanradiohistory.com 368 TV and Broadcast Interference THE SHIELD BOX CI (SNORT 0 300-OHM TO LINE FROM INPUT S ANTENNA OF TV I SE LEADS 300 OHM LINE FROM ANTENNA _ L2 pA FOR COAx FIiTING Jl ANTENNA TERMINALS OF TV SET TO Figure TUNED TRAPS FOR THE TRANSMITTER FUNDAMENTAL The arrangement at (A) has proven to be effective in eliminating the condition of general blocking as caused by a 28 -Mc. transmitter in the vicinity of a TV receiver. The tuned circuits LI -CI are resonated separately to the frequency of transmission. The adjustment may be done at the station, or it may be accomplished at the TV receiver by tuning for minimum interference on the TV screen. FOR 50 -75 Shown at (B) is an alternative arrangement a series -tuned circuit across the anten- naced as a source of interference to reception, work may then be begun on or in the vicinity of the transmitter toward eliminating the other two types of interference. less standard BCItype practice is most commonly used in t a k i n g out fundamental interference. Wavetraps and filters are installed, and the antenna system may or may not be modified so as to offer less response to the signal from the amateur transmitter. In regard to a comparison between wavetraps and filters, the same considerations apply as have been effective in regard to BCI for many years; wavetraps are quite effective when properly installed and adjusted, but they Taking Out the Fundamental More or TERM ON TV SET I COAX FITTING 3 OHM COAXIAL LINE Figure 1 na terminals of the TV set. The tuned circuit should be resonated to the operating frequency of the transmitter. This arrangement gives less attenuation of the interfering signal than that at (A); the circuit has proven effective with interference from transmitters on the 50 -Mc. band, and with low power 28 -Mc. transmitters. ANTENNA C2 II-E--d --F L3 © Li ` TO 300 -OHM LINE, SHIELDED OR UNSHIELDED C2 TO TV ANTENNA with RADIO 2 HIGH -PASS TRANSMISSION LINE FILTERS The arrangement at (A) will stop the passing of all signals below about 45 Mc. from the antenna transmission line into the TV set. Coils LI ore each 1.2 microhenrys (17 turns no. 24 enam. closewound on 1/2-inch dia. polystyrene rod) with the center tap grounded. It will be found best to scrape, twist, and solder the center tap before winding the coil. The number of turns each side of the tap may then be varied until the tap is in the exact center of the winding. Coil L2 is 0.6 microhenry (12 turns no. 24 enam. closewound on 1/2-inch dia. polystyrene rod). The capacitors should be about 16.5 Wald., but either 15 or 20 pµtd. ceramic capacitors will give satis- factory results. A similar filter for coaxial antenna transmission line is shown at (B). Both coils should be 0.12 microhenry (7 turns no. 18 enam. spaced to % inch on 1/2-inch dia. polystyrene rod). Capacitors C2 should be 75 µµtd. midget ceramics, while C3 should be a 40 -µµtd. ceramic. must be readjusted whenever the band of operation is changed, or even when moving from one extreme end of a band to the other. Hence, wavetraps are not recommended except when operation will be confined to a relatively narrow portion of one amateur band. However, figure 1 shows two of the most common signal trapping arrangements. High -Pass Filters High -pass filters in the antenna lead of the TV set have proven to be quite sat i s factory as a means of eliminating TVI of the overloading type. In many cases when the interfering transmitter is operated only on the bands below 7.3 Mc., the use of a high -pass filter in the antenna lead has completely eliminated all www.americanradiohistory.com HANDBOOK Harmonic Radiation 369 TVI. In some cases the installation of a high pass filter in the antenna transmission line and an a -c line filter of a standard variety has proven to be completely effective in eliminating the interference from a transmitter operat- ing in one of the lower bands. In frequency amateur general, it is suggested that commercial- ly manufactured high-pass filters be purchased. Such units are available from a number of manufacturers at a relatively moderate cost. However, such units may be home constructed; suggested designs are given in figures 2 and 3. Types for use both with coaxial and with balanced transmission lines have been shown. In most cases the filters may be constructed in one of the small shield boxes which are now on the market. Input and output terminals may be standard connectors, or the inexpensive type of terminal strips usually used on BC and TV sets may be employed. Coaxial terminals should of course be employed when a coaxial feed line is used to the antenna. In any event the leads from the filter box to the TV set should be very short, including both the antenna lead and the ground lead to the box itself. If the leads from the box to the set have much length, they may pick up enough signal to nullify the effects of the high -pass filter. Blocking from 50-Me. Signals Operation on the 50 -Mc. amateur band in an area where channel 2 is in use for TV imposes a special problem in the matter of blocking. The input circuits of most TV sets are sufficiently broad so that an amateur signal on the 50 -Mc. band will ride through with little attenuation. Also, the normal TV antenna will have a quite large response to a signal in the 50 -Mc. band since the lower limit of channel 2 is 54 Mc. High -pass filters of the normal type simply are not capable of giving sufficient attenuation to a signal whose frequency is so close to the necessary pass band of the filter. Hence, a resonant circuit element, as illustrated in figure :, must be used to trap out the amateur field at the input of the TV set. The trap must be tuned or the section of transmission line cut, if a section of line is to be used for a particular frequency in the 50 -Mc. band. This frequency will have to be near the lower frequency limit of the 50 -Mc. band to obtain adequate rejection of the amateur signal while still not materially affecting the response of the receiver to channel 2. Elimination of Spurious Emissions All spurious e m i s s i o n s from amateur transmitters (ignoring harmonic signals eliminated to corn- for the time being) must be Figure 3 SERIES -DERIVED HIGH -PASS FILTER This filter is designed for use in the 300 -ohm transmission line from the TV antenna to the TV receiver. Nominal cutoff frequency is 36 Mc. and maximum rejection is at about 29 Mc. Ct,C6- 15 -µµfd. zero- coefficient ceramic C2,C3,C4,C5-20 -i fd. zero -coefficint cramic L1,Lt -2.0 µh. About 24 turns no. 28 d.c.c. wound to ?éi on '4" diameter polystyrene rod. Turns should be adjusted until the coil resonates to 29 Mc. with the asociated 15- ptµfd. capacitor. L2-0.66 i h., 14 turns no. 28 d.c.c. wound to Se" on t/ " die. polystyrene rod. Adjust turns to resonate externally to 20 Mc. with an auxiliary 100- µ{tfd. capacitor whose value is accurately known. ply with FCC r e g u l a t i o n s. But in the past many amateur transmitters have emitted spurious signals as a result of key clicks, parasities, and overmodulation transients. In most cases the operators of the transmitters were not aware of these emissions since they were radiated only for a short distance and hence were not brought to his attention. But with one or more TV sets in the neighborhood it is probable that such spurious signals will be brought quickly to his attention. 19 -2 Harmonic Radiation After any condition of blocking at the TV receiver has been eliminated, and when the transmitter is completely free of transients and parasitic oscillations, it is probable that TVI will be eliminated in certain cases. Certainly general interference should be eliminated, particularly if the transmitter is a well designed affair operated on one of the lower frequency bands, and the station is in a high signal TV area. But when the transmitter is to be operated on one of the higher frequency bands, and particularly in a marginal TV area, the job of TVI -proofing will just have begun. The elimination of harmonic radiation from the transmitter is a difficult and tedious job which must be done in an orderly manner if completely satisfactory results are to be obtained. www.americanradiohistory.com 370 THE TV and Broadcast Interference TRANSMITTER FUNDAMENTAL 7.0 7 4TH 3RD 2ND 42-44 TV I.F 3 8TH 7TH 6TH 5TH 21 -21 9 CHA NEL 2 NEW 14.0 2 -43 14!4 FM CHANNEL CHANNEL CHANNEL NEW pd BROAD TV 1. F. 02 CAST 63 -64 35 64 -65 6 105' 110725 cHt5NEL V1 271 53.92- 54 46 23 60.66- 169.193 FM CHAfNNEL V F5) 26.96 ELC EL © 21.0 ( 70-73 7 84 -664 96-100.6 70-72 56 -57.6 107H 9TH 56-56 4 63-65 701F RADIO 9 169 107.64- CHANNEL i (J 2C 28.0 56 -59 4 291 7 CHANNEL CHANNEL 50.0 100-105 5410 BROAD- FM 166-1762199-2079 B4 -89.1 O 5 EL 0 216 10692 61 69 CH.jtJNEL 210-214 S C BROAD- CHANNEL 1,1E}á I`11, 1 50 -486 200.216 500-540 HAy ST 3 POSSIBLE INTERFERENCE 1 TO Figure U-H-F CHANNELS 4 HARMONICS OF THE AMATEUR BANDS Shown are the harmonic frequency ranges of the amateur bands between 7 and 54 Mc., with the TV channels (and TV i -f systems) which are most likely to receive interference from these harmonics. Under certain conditions amateur signals in the 1.8 and 3.5 Mc. bands con cause interference as a result of direct pickup in the video systems of TV receivers which are not ade- quately shielded. cross -hatch or herringbone pattern on the TV screen. This same general type of picture also will occur in the case of a narrow -band FM signal either with or First it is well to become familiar with the TV channels presently assigned, with the TV intermediate frequencies commonly used, and with the channels which will receive interference from harmonics of the various amateur bands. Figures 4 and 5 give this information. Even a short inspection of figures 4 and 5 will make obvious the seriousness of the interference which can be caused by harmonics of amateur signals in the higher frequency bands. With any sort of reasonable precautions in the design and shielding of the transmitter it is not likely that harmonics higher than the 6th will be encountered. hence the main offenders in the way of harmonic interference will be those bands above 14 -Mc. Nature of Harmonic Interference Investigations into the nature of the interference caused by amateur signals on the TV screen, assuming that blocking has been eliminated as described earlier in this chapter, have revealed the following facts: An unmodulated carrier, such as a c -w 1. signal with the key down or an AM signal without m o d u l at ion, will give a without modulation. A relatively strong AM signal will give in addition to the herringbone a very serious succession of light and dark bands across the TV picture. 3. A moderate strength c -w signal without transients, in the absence of overloading of the TV set, will result merely in the turning on and off of the herringbone on the picture. To discuss condition (1) above, the herringbone is a result of the beat note between the TV video carrier and the amateur harmonic. Hence the higher the beat note the less obvious will be the resulting cross -hatch. Further, it has been shown that a much stronger signal is required to produce a discernible herringbone when the interfering harmonic is as far away as possible from the video carrier, without running into the sound carrier. Thus, as a last resort, or to eliminate the last vestige of interference after all corrective measures have been taken, operate the transmitter on a frequency such that the interfer2. www.americanradiohistory.com HANDBOOK Harmonic VIDEO SOUND U u O P 1 I - TV 'CHANNEL' 1 1 1 371 00 t TV %/=ii /.% %/ 1 1 1 TV 'CHANNEL I I Interference I I I TV 'CHANNEL I O I I I I 76 I CASTO , I\\ O I V 1 IR \\ t TV 'CHANNEL O I t e2 88 foe LOW BAND VIDEO SOUND ro ü it r I TV 1 ICHANNELI I® I I 1 ú I 1 TV 1 I ICHANNELI I O I I 1 1 I TV I I I O TV 1 ICHANNELI I I CHANNEL I I I @ 1 t I 1 i 7 TV 1CHANNELI I I I 1 1 1 I ICHANNEL I I ® I 192 198 TV ICHANNELI 1 1 1 1 1 204 210 ï T I I 1 Y TV I I I I 0 I II 216 HIGH BAND Figure 5 FREQUENCIES OF THE V -H -F TV CHANNELS Showing the frequency ranges of TV channels 2 through 13, with the picture carrier and sound carrier frequencies also shown. ing harmonic will fall as far as possible from the picture carrier. The worst possible interference to the picture from a continuous carrier will be obtained when the interfering signal is very close in frequency to the video carrier. Isolating Throughout the testing procedure it will be necessary to have some sort of indicating device as a means of determining harmonic field intensities. The best indicator for field intensities some distance from the transmitting antenna will probably be the TV receiver of some neighbor with whom friendly relations are still maintained. This person will then be able to give a check, occasionally, on the relative nature of the interference. But it will probably be necessary to go and check yourself periodically the results obtained, since the neighbor probably will not be able to give any sort of a quantitative analysis of the progress which has been made. An additional device for checking relatively high field intensities in the vicinity of the transmitter will be almost a necessity. A simple crystal diode wavemeter, shown in figure 6 will accomplish this function. Also, it will be very helpful to have a receiver, with an S meter, capable of covering at least the 50 to 100 Mc. range and preferably the range to 216 Mc. This device may consist merely of the station receiver and a simple converter using the two halves of a 6J6 as oscillator and mixer. the Source of the Interference The first check can best be made with the neighbor who is receiving the most serious or the most general interference. Turn on the transmitter and check all channels to determine the extent of the interference and the number of channels affected. Then disconnect the antenna and substitute a group of 100 -watt lamps as a dummy load for the transmitter. Experience has shown that 8 100 -watt lamps connected in two seriesed groups of four in parallel will take the output of a kilowatt transmitter on 28 Mc. if connections are made symmetrically to the group of lamps. Then note the interference. Now remove plate voltage from the final amplifier and determine the extent of interference caused by the exciter stages. In the average case, when the final amplifier is a beam tetrode stage and the exciter is . 10' PICKUP WIRE 5T X0.3 DIA.18E, 0.5 LONG COVERAGE -35-140 MC. IN 34 Figure 6 Crystal -diode wavemeter suitable for checking high-intensity harmonics in TV region. www.americanradiohistory.com 372 relatively low powered and adequately shielded, it will be found that the interference drops materially when the antenna is removed and a dummy load substituted. It will also be found in such an average case that the interference will stop when the exciter only is operating. RADIO THE TV and Broadcast Interference Lx L4 L3 L5 Lip l T It should be made clear at this point that the l e v e l of power used at the transmitter is not of great significance in the basic harmonic reduction problem. The difference in power level between a 20 -watt transmitter and one rated at a kilowatt is only a matter of about 17 db. Yet the degree of harmonic attenuation required to eliminate interference caused by harmonic radiation is from 80 to 120 db, depending upon the TV signal strength in the vicinity. This is not to say that it is not a simpler job to eliminate harmonic interference from a low -power transmitter than from a kilowatt equipment. It is simpler to suppress harmonic radiation from a low -power transmitter simply because it is a much easier problem to shield a low -power unit, and the filters for the leads which enter the transmitter enclosure may be constructed less expensively and smaller for a low -power unit. Ca C3 Cz Transmitter Power Level 19 -3 Figure 7 FILTER SCHEMATIC DIAGRAMS LOW -PASS The filter illustrated at (A) uses mderived terminating half sections at each end, with three constant -k mid -sections. The filter at (B) is essentially the same except that the center section has been changed to act as on m- derived section which can be designed to offer maximum attenuation to channels 2, 4, 5, or 6 in accordance with the constants given below. Cutoff frequency is 45 Mc. in all cases. All coils, except L4 in (B) above, are wound 1/2 "i.d. with 8 turns per inch. The (A) Filter C,,C6 -41.5µµ4d. (40 µµfd. will be found suit- Low -Pass Filters able.) After the transmitter has been shielded, and all power leads have been filtered in such a manner that the transmitter shielding has not been rendered ineffective, the only remaining available exit for harmonic energy lies in the antenna transmission line. Hence the main burden of harmonic attenuation will fall on the filter installed between the output of the transmitter and the antenna system. Experience has shown that the low -pass filter can best be installed externally to the main transmitter enclosure, and that the transmission line from the transmitter to the low pass filter should be of the coaxial type. Hence the majority of low -pass filters are designed for a characteristic impedance of 52 ohms, so that RG -8 /U cable (or RG -58/U for a small transmitter) may be used between the output of the transmitter and the antenna transmission line or the antenna tuner. Transmitting -type low -pass filters for amateur use usually are designed in such a manner as to pass frequencies up to about 30 Mc. without attenuation. The nominal cutoff frequency of the filters is usually between 38 and 45 Mc., and m- derived sections with maximum attenuation in channel 2 usually are included. Well- designed filters capable of carrying any power level up to one kilowatt are low -pass C2, C3, C4-136 µµfd. (130 to 140 µµid. may be used.) -0.2 ph; 3 S t. no. 14 L6 L2, L4-0.3 ph; 5 t. no. 12 L3, L4, -0.37 ph; 615 t. no. 12 L,, The (B) Filter with Mid- Section tuned to Channel 2 (58 Mc..) C C6 C2, Ca C3 -41.5 -136 -87 pfd. µpfd. (50 µpfd. fixed and 75 able in parallel.) L3, L7-0.2 ph; L2, L3, L3 , L6 3 Si L4-0.09 ph; t. no. 14 55" die. by -0.3 2 µpfd. vari- t. no. 14 ph; 5 t. no. 12 '4 " long The (B) Filter with Mid -Sction tuned to Channel 4 (71 Mc.). All components same except that: -106 -0.33 ph; 6 t. L4 -0.05 ph; 1)5 t. C3 L3, L3 long. no. 12 no. 14, 3 '8 " dia. by 3/8 " The (B) Filter with Mid -Section tuned to Channel 5 (81 Mc.). Change the following: C3 -113 µpfd. L,, L4-0.34 ph; 6 t. no. 12 L4-0.033 ph; 1 t. no. 14 3/8" dia. The (B) Filter with Mid -Section tuned to Channel 6 (86 Mc.). All comp is are essentially the same except that the theoretical value of La Is changed to 0.03 ph., and the capacitance of C3 is changed to 117 µpfd. www.americanradiohistory.com HANDBOOK Low available commercially from several manufacturers. Alternatively, filters in kit form are available from several manufacturers at a somewhat lower price. Effective filters may be home constructed, if the test equipment is available and if sufficient care is taken in the construction of the assembly. Figures 7, 8 and 9 illustrate high- performance low-pass filters which are suitable for home construction. All are constructed in slip -cover aluminum boxes (ICA no. 29110) with dimensions of 17 by 3 by 2% inches. Five aluminum baffle plates have been installed in the chassis to make six shielded sections within the enclosure. Feed-through bushings between the shielded sections are Johnson no. 135 -55. Both the (A) and (B) f i t e r types are designed for a nominal cut -off frequency of 45 Mc., with a frequency of maximum rejection at about 57 Mc. as established by the terminating half- sections at each end. Characteristic impedance is 52 ohms in all cases. The alternative filter designs diagrammed in figure 7B have provision for an additional rejection trap in the center of the filter unit which may be designed to offer maximum r ejection in channel 2, 4, 5, or 6, depending upon which channel is likely to be received in the area in question. The only components which must be changed when changing the frequency of the maximum rejection notch in the center of the filter unit are inductors L4, and L,, and capacitor C,. A trimmer capacitor has been included as a portion of C, so that the frequency of maximum rejection can be tuned accurately to the desired value. Reference to figures 5 and 6 will show the amateur bands which are Pass Filters 373 .I Construction of Low -Pass Filters 1 L Figure 8 PHOTOGRAPH OF THE (B) FILTER WITH THE COVER IN PLACE most likely to cause interference to specific TV channels. Either high -power or low -power components may be used in the filters diagrammed in figure 7. With the small Centralab TCZ zero- coefficient ceramic capacitors used in the filter units of figure 7A or figure 7B, power levels up to 200 watts output may be used without danger of damage to the capacitors, provided the filter is feeding a 52 -ohm resistive load. It may be practicable to use higher levels of power with this type of ceramic capacitor in the filter, but at a power level of 200 watts on the 28 -Mc. band the capacitors run just perceptibly warm to the touch. As a point of interest, it is the current rating which is of significance to the capacitors used in filter s such as illustrated. Since current ratings for small capacitors such as these are not readily available, it is not possible to establish an accurate power rating for such a unit. The high -power unit illustrated in figure 9, which uses Centralab type 850S and 854S capacitors, Figure 9 PHOTOGRAPH OF THE (B) FILTER WITH COVER REMOVED The mid -section in this filter is adjusted for maximum rejection of channel 4. Note that the main coils of the filter are mounted at on angle of about 45 degrees so that there will be minimum inductive coupling from one section to the next through the holes in the aluminum partitions. Mounting the coils in this manner was found to give a measurable improvement in the attenuation characteristics of the filter. www.americanradiohistory.com 374 THE TV and Broadcast Interference has proven quite suitable for power levels up to one kilowatt. Capacitors C2, C4, and C, can be standard manufactured units with normal 5 per cent tolerance. The coils for the end sections can be wound to the dimensions given (Lt, L6, and L,). Then the resonant frequency of the series resonant end sections should be checked with a grid -dip meter, after the adjacent input or output terminal has been shorted with a very short lead. The coils should be squeezed or spread until resonance occurs at 57 Mc. The intermediate m- derived section in the filter of figure 7B may also be checked with a grid -dip meter for resonance at the correct rejection frequency, after the hot end of L4 has been temporarily grounded with a low- inductance lead. The variable capacitor portion of C, can be tuned until resonance at the correct frequency has been obtained. Note that there is so little difference between the constants of this intermediate section for channels 5 and 6 that variation in the setting of C, will tune to either channel without materially changing the operation of the filter. The coils in the intermediate sections of the filter (Lr, L4, and L3 in figure 7A, and L2, L3, L4, and L6 in figure 7B) may be checked most conveniently outside the filter unit with the aid of a small ceramic capacitor of known value and a grid -dip meter. The ceramic capacitor is paralleled a c r o s s the small coil with the shortest possible leads. Then the assembly is placed atop a cardboard box and the RADIO C L resonant frequency checked with a grid -dip meter. A Shure reactance slide rule may be used to ascertain the correct resonant frequency for the desired L -C combination and the coil altered until the desired resonant frequency is attained. The coil may then be installed in the filter unit, making sure that it is not squeezed or compressed as it is being installed. However, if the coils are wound exactly as given under figure 10, the filter may be assembled with reasonable assurance that it will operate as designed. filter low -pass con nected in the output transmission line of the transmitter is capable of affording an enormous degree of harmonic attenuation. However, the filter must be operated in the correct manner or the results obtained will not be up to expectations. In the first place, all direct radiation from the transmitter and its control and power leads must be suppressed. This subject has been discussed in the previous section. Secondly, the filter must be operated into a load impedance approximately equal to its design characteristic impedance. The filter itself will Using Low -Pass Filters The Figure 10 SCHEMATIC OF THE SINGLE -SECTION HALF -WAVE FILTER The constants given below are for a characteristic impedance of 52 ohms, for use with RG -8 /U and RG -58/1/ cable. Coil Lt should be checked for resonance at the operating frequency with Cr, and the same with L2 and C4. This check can be mode by soldering a low- inductance grounding strap to the lead between L1 and L2 where it passes through the shield. When the coils have been trimmed to resonance with a grid -dip meter, the grounding strap should of course be removed. This filter type will give an attenuation of about 30 db to the second harmonic, about 48 db to the third, about 60 db to the fourth, fifth, and so on increasing at a rate of about 30 db per octave. Ct,C2,C3,C4- Silver mica or small ceramic for low power, transmitting type ceramic for high power. Capacitance for different bands is given below: 160 meters -1700 µµfd. 67 to the -850 -440 -220 meters -110 80 meters 40 meters 20 meters µµfd. µµfd. µµid. µµfd. µµfd. 10 6 meters be made up of -60 L, -May sections of B&W Mini ductor for power levels below 250 watts, or of 12 to one kilowatt. Apfor power up no. enom. proximate dimensions for the coils are given below, but the coils should be trimmed to resonate at the proper frequency with a grid -dip meter as discussed above. All coils except the ones for 160 meters are wound 8 turns per Inch. 160 meters -4.2 µh; 22 turns no. 16 enam., dia. 2" long 80 meters -2.1 µh; 13 t. 1" dia. (No. 3014 Mini ductor or no. 12) 40 meters -1.1 µh; 8 t. 1" dla. (No. 3014 or no. L 1 12 at 8 t.p.i.) 20 meters -0.55 µh; 7 no. 12 at 8 t.p.i.) t',l" dia. (No. 3010 or meters -0.3 µh; 6 t. S4" dia. (No. 3002 or no. 12 of 8 6 meters -0.17 µh; 4 t. 14" dia. (No. 3002 or no. 12 at 8 t.p.i.) 10 tpi.) have very low losses (usually less than 0.5 db) when operated into its nominal value of resistive load. But if the filter is mis- terminated its losses will become excessive, and it will not present the correct value of load impedance to the transmitter. If a f i l t e r, being fed from a high -power transmitter, is operated into an incorrect termination it may be damaged; the coils may be overheated and the capacitors destroyed as a result of excessive r -f currents. Hence it is wise, when first installing a low -pass filter, www.americanradiohistory.com HANDBOOK Figure Broadcast Interference 375 11 HALF -WAVE FILTER FOR THE 28 -MC. BAND Showing one possible type of construction of o 52 -ohm half -wave filter for relatively low power operation on the 28 -Mc. bond. to check the standing -wave ratio of the load being presented to the output of the filter with a standing -wave meter of any of the conventional types. Then the antenna termination or the antenna coupled should be adjusted, with low power on the transmitter, until the s.w.r. of the load being presented to the filter is less than 2.0, and preferably below 1.5. Half -wave filters ( "Harmonikers") have been discussed in various publications including the Nov. -Dec. 1949 GE Ham News. Such filters are relatively simple and offer the advantage that they present the same value of impedance at their input terminals as appears as load across their output terminals. Such filters normally are used as one-band affairs, and they offer high attenuation only to the third and higher harmonics. Design data on the half wave filter is given in figure 10. Construction of half-wave filters is illustrated in figure 11. Half -Wove Filters 19 -4 Broadcast Interference Interference to the reception of signals in the broadcast band (540 to 1600 kc.) or in the FM broadcast band (88 to 108 Mc.) by amateur transmissions is a serious matter to those amateurs living in densely populated areas. Although broadcast interference has recently been overshadowed by the seriousness of television interference, the condition of BCI is still present. In general, signals from a transmitter operating properly are not picked up by receivers tuned to other frequencies unless the receiver is of inferior design, or is in poor condition. Therefore, if the receiver is of good design and is in good repair, the burden of rectifying the trouble rests with the owner of the interfering station. Phone and c -w stations both are capable of causing broadcast interference, key -click annoyance from the code transmitters being particularly objectionable. A knowledge of each of the several types of broadcast interference, their cause, and methods of eliminating them is necessary for the successful disposition of this trouble. An effective method of combating one variety of interference is often of no value whatever in the correction of another type. Broadcast interference seldom can be cured by "rule of thumb" procedure. Broadcast interference, as covered in this section refers primarily to standard (amplitude modulated, 550 -1600 kc.) broadcast. Interference with FM broadcast reception is much less common, due to the wide separation in frequency between the FM broadcast band and the more popular amateur bands, and due also to the limiting action which exists in all types of FM receivers. Occasional interference with FM broadcast by a harmonic of an amateur transmitter has been reported; if this condition is encountered, it may be eliminated by the procedures discussed in the first portion of this chapter under Television Interference. The use of frequency -modulation transmission by an amateur station is likely to result in much less interference to broadcast reception than either amplitude-modulated telephony or straight keyed c.w. This is true because, insofar as the broadcast receiver is concerned, the amateur FM transmission will consist of a plain unmodulated carrier. There will be no key clicks or voice reception picked up by the b-c-1 set (unless it happens to be an FM receiver which might pick up a harmonic of the signal), although there might be a slight click when the transmitter is put on or taken www.americanradiohistory.com 376 THE TV and Broadcast Interference Figure RADIO 13 HIGH -ATTENUATION WAVE -TRAP CIRCUIT circuits may be tuned to the same frequency for highest attenuation of a strong signal, or the two traps may be tuned separately for different bands of operation. The two Figure 12 WAVE -TRAP CIRCUITS The circuit at (A) is the most common arrangement, but the circuit at (B) may give improved results under certain conditions. Manufactured wave traps for the desired band of operation may be purchased or the traps may be assembled from the data given in figure 14. off the air. This is one reason why narrow band FM has become so popular with phone enthusiasts who reside in densely populated areas. Interference Depending upon whether it is traceable directly to causes within the station or within the receiver, broadcast interference may be divided into two main classes. For example, that type of interference due to transmitter over -modulation is at once listed as b e i n g caused by improper operation, while an interfering signal that tunes in and out with a broadcast station is probably an indication of cross modulation or image response in the receiver, and the poorly- designed input stage of the receiver is held liable. The various types of interference and recommended cures will be discussed in the following paragraphs. Classifications This is not a tunable effect, but a total blocking of the receiver. A more or less complete "washout" covers the entire receiver range when the carrier is switched on. This produces either a complete blotting out of all broadcast stations, or else knocks down their volume several decibels depending upon the severity of the interference. Voice modulation of the carrier causing the blanketing will be highly distorted or even Blanketing unintelligible. Keying of the carrier which produces the blanketing will cause an annoying fluctuation in the volume of the broadcast signals. Blanketing generally occurs in the immediate neighborhood (inductive field) of a powerful transmitter, the affected area being directly proportional to the power of the transmitter. Also it is more prevalent with transmitters which operate in the 160 -meter and 80 -meter bands, as compared to those on the higher frequencies. The remedies are to (1) shorten the receiving antenna and thereby shift its resonant frequency, or (2) remove it to the interior of the building, (3) change the direction of either the receiving or transmitting antenna to minimize their mutual coupling, or (4) keep the interfering signal from entering the receiver input circuit by installing a wavetrap tuned to the signal frequency (see figure 12) or a low-pass filter as shown in figure 21. A suitable wave -trap is quite simple in construction, consisting only of a coil and midget variable capacitor. When the trap circuit is tuned to the frequency of the interfering signal, little of the interfering voltage reaches the grid of the first tube. Commercially manufactured wave -traps are available from several concerns, including the J. W. Miller Co. in Los Angeles. However, the majority of amateurs prefer to construct the traps from spare components selected from the "junk box." The circuit shown in figure 13 is particularly effective because it consists of two traps. The shunt trap blocks or rejects the frequency to which it is tuned, while the series trap across the antenna and ground terminals of the receiver provides a very low impedance path to ground at the frequency to www.americanradiohistory.com Wavetraps HANDBOOK BAND COIL, CAPACITOR, L C 1.8 Mc. Inch no. 30 enom. 75 -,..ofd. var. 1 closewound on 1" form 3.5 Mc. 42 turns no. 30 enam. 50 -cAfd. var. closewound on 1" form 7.0 Mc. 23 turns no. 24 enam. 50 -µsfd. var. closewound on 1" form 14 Mc. 10 turns no. 24 enam. 50 -Aµfd. var. closewound on 1" form 21 Mc. 7 28 Mc. 4 50 Mc. 3 turns no. 24 enam. 50 -ccfd. var. closewound on 1" form turns no. 24 enam. 25 -µµfd. vor. closewound on 1" form t spaced I no. 24 cram. on 1" form /2" Figure 25 -Acfd. Figure 15 MODIFICATION OF THE FIGURE CIRCUIT var. circuit arrangement the paralll -tuned tank is inductively coupled to the antenna lead with a 3 to 6 turn link instead of being placed directly in series with the antenna lead. which it is tuned and by- passes the signal to ground. In moderate interference cases, either the shunt or series trap may be used alone, while similarly, one trap may be tuned to one of the frequencies of the interfering transmitter and the other trap to a different interfering frequency. In either case, each trap is effective over but a small frequency range and must be readjusted for other frequencies. The wave -trap must be installed as close to the receiver antenna terminal as practicable, hence it should be as small in size as possible. The variable capacitor may be a midget air -tuned trimmer type, and the coil may be wound on a 1 -inch dia. form. The table of figure 14 gives winding data for wave -traps built around standard variable capacitors. For best results, both a shunt and a series trap should be employed as shown. Figure 15 shows a two- circuit coup e d wave -trap that is somewhat sharper in tuning and more efficacious. The specifications for the secondary coil L, may be obtained from the table of figure 14. The primary coil of the shunt trap consists of 3 to 5 closewound turns of the same size wire wound in the same direction on the same form as L, and separated 1 latter by Overmodulation A '4 13 In this 14 COIL AND CAPACITOR TABLE FOR AMATEUR -BAND WAVETRAPS from the 377 of an inch. carrier modulated in excess of 100 per cent acquires sharp cutoff periods which give rise to transients. These transients create a broad signal and generate spurious responses. Transients caused by overmodulation of a radio -telephone signal may at the same time bring about impact or shock excitation of nearby receiving antennas and power lines, generating interfering signals in that manner. Broadcast interference due to overmodulation is frequently encountered. The remedy is to reduce the modulation percentage or to use a clipper -filter system or a high -level splatter suppressor in the speech circuit of the transmitter. Cross modulation or cross talk is characterized by the amateur signal riding in on top of a strong broadcast signal. There is usually no heterodyne note, the amateur signal being tuned in and out with the program carriers. This effect is due frequently to a faulty input stage in the affected receiver. Modulation of the interfering carrier will swing the operating point of the input tube. This type of trouble is seldom experienced when a variable-µ tube is used in the input stage. Where the receiver is too ancient to incorporate such a tube, and is probably poorly shielded at the same time, it will be better to attach a wave -trap of the type shown in figure 12 rather than to attempt rebuilding of the receiver. The addition of a good ground and a shield can over the input tube often adds to the effectiveness of the wave -trap. Cross Modulation Transmission via Capacitive Coupling A small amount of ca- pacitive coupling is now widely used in receiver r.f. and antenna transformers as a gain booster at the high- frequency end of the tuning range. The coupling capacitance is obtained by means of a small loop of wire cemented close to the grid end of the secondary winding, with one end directly connected to the plate or antenna end of the primary winding. (See figure 16.) www.americanradiohistory.com 378 THE TV and Broadcast Interference CAPACITIVE COUPLING LOOP Figure 16 CAPACITIVE BOOST COUPLING CIRCUIT Such circuits, included within the broadcast receiver to bring up the stage gain at the high -frequency end of the tuning ronge, have a tendency to increase the susceptibility of the receiver to interference from amateur band transmissions. It is easily seen that a small capacitor at this position will favor the coupling of the higher frequencies. This type of capacitive coupling in the receiver coils will tend to pass amateur high- frequency signals into a receiver tuned to broadcast frequencies. The amount of capacitive coupling may be reduced to eliminate interference by moving the coupling turn further away from the secondary coil. However, a simple wave -trap of the type shown in figure 12, inserted at the antenna input terminal, will generally accomplish the same result and is more to be recommended than reducing the amount of capacitive coupling (which lowers the receiver gain at the high- frequency end of the broadcast band). Should the wave -trap alone not suffice, it will be necessary to resort to a reduction in the coupling capacitance. In some simple broadcast receivers, capacitive coupling is obtained by closely coupled primary and secondary coils, or as a result of running a long primary or antenna lead close to the secondary coil of an unshielded antenna coupler. strong local carriers applied to a non -linear impedance, the beat note resulting from cross -modulation between them may fall on some frequency within the broadcast band and will be audible at that point. If such a "phantom" signal falls on a local broadcast frequency, there will be heterodyne interference as well. This is a common occurence with broadcast receivers in the neighborhood of two amateur stations, or an amateur and a police station. It also sometimes occurs when only one of the stations is located in the immediate vicinity. Phantoms With two RADIO As an example: an amateur signal on 3514 kc. might beat with a local 2414 -kc. police carrier to produce a 1100 -kc. phantom. If the two carriers are strong enough in the vicinity of a circuit which can cause rectification, the 1100 -kc. phantom will be heard in the broadcast band. A poor contact between two oxidized wires can produce rectification. Two stations must be transmitting simultaneously to produce a phantom signal; when either station goes off the air the phantom disappears. Hence, this type of interference is apt to be reported as highly intermittent and might be difficult to duplicate unless a test oscillator is used "on location" to simulate the missing station. Such interference cannot be remedied at the transmitter, and often the rectification takes place some distance from the receivers. In such occurrences it is most difficult to locate the source of the trouble. It will also be apparent that a phantom might fall on the intermediate frequency of a simple superhet receiver and cause interference of the untunable variety if the manufacturer has not provided an i -f wave -trap in the antenna circuit. This particular type of phantom may, in addition to causing i-f interference, generate harmonics which may be tuned in and out with heterodyne whistles from one end of the receiver dial to the other. It is in this manner that birdies often result from the operation of nearby amateur stations. G hen one component of a phantom is a steady, unmodulated carrier, only the intelligence present on the other carrier is conveyed to the broadcast receiver. Phantom signals almost always may be identified by the suddenness with which they are interrupted, signalizing withdrawal of one party to the union. This is especially baffling to the inexperienced interference -locater, who observes that the interference suddenly disappears, even though his own transmitter remains in operation. If the mixing or rectification is taking place in the receiver itself, a phantom signal may be eliminated by removing either one of the contributing signals from the receiver input circuit. A wave -trap of the type shown in figure 12, tuned to either signal, will do t h e trick. If the rectification is taking place outside the receiver, the wave -trap should be tuned to the frequency of the phantom, instead of to one of its components. I -f wave -traps may be built around a 2.5- millihenry r -f choke as the inductor, and a compression -type mica padding capacitor. The capacitor should have a capacitance range of 250 -525 µµfd. for the 175- and 206 -kc. intermediate frequencies; 65 -175 µµfd. for 260 -kc. and other intermedi- www.americanradiohistory.com HANDBOOK Audio ates lying between 250- and 400 -kc; and 17 -80 µµEd. for 456 -, 465-, 495 -, and 500 -kc. Slightly more capacitance will be required for resonance with a 2.1 millihenry choke. This sort of interference arises from the transmitter itself. The radiation of any signal (other than the intended carrier frequency) by an amateur station is prohibited by FCC regulations. Spurious radiation may be traced to imperfect neutralization, parasitic oscillations in the r -f or modulator stage s, or to "broadcast- band" variable- frequency oscillators or e.c.o.'s. Low- frequency parasitics may actually occur on broadcast frequencies or their near sub harmonics, causing direct interference to programs. An all -wave monitor operated in the vicinity of the transmitter will detect these spurious signals. The remedy will be obvious in individual cases. Elsewhere in this book are discussed methods of complete neutralization and the suppression of parasitic oscillations in r -f and audio stages. Rectification 379 HIGH -MU TUBE SUCH AS 12507 Spurious Emissions Inexpensive tab e- model a -c /d -c receivers are particularly susceptible to interference from amateur transmissions. In fact, it may be said with a fair degree of assurance that the majority of BCI encountered by amateurs operating in the 1.8 -Mc. to 29-Mc. range is a result of these inexpensive receivers. In most cases the receivers are at fault; but this does not absolve the amateur of his responsibility in attempting to eliminate the interference. A -c /d -e Receivers 1 cases of interference receivers, par- Stray Receiver In most Rectification to inexpensive ticularly those of the a -c /d -c type, it will be found that stray receiver rectification is causing the trouble. The offending stage usually will be found to be a high -mu triode as the first audio stage following the second detector. Tubes of this type are quite non-linear in their grid characteristic, and hence will readily rectify any r -f signal appearing between grid and cathode. The r -f signal may get to the tube as a result of direct signal pickup due to the lack of shielding, but more commonly will be fed to the tube from the power line as a result of the series heater string. The remedy for this condition is simply to insure that the cathode and grid of the high -mu audio tube (usually a 12SQ7 or equivalent) are at the same r-f potential. This is accomplished by placing an r-f by -pass capacitor with the shortest possible leads directly from grid to cathode, and then adding an impedance in the lead from the volume control to the grid of the Figure 17 CIRCUITS FOR ELIMINATING AUDIO - STAGE RECTIFICATION audio tube. The impedance may be an amateur band r -f choke (such as a National R -I00U) for best results, but for a majority of cases it will be found that a 47,000 -ohm V2-watt resistor in series with this lead will giv.e satisfactory operation. Suitable circuits for such an operation on the receiver are given in figure 17. In many a.c. -d.c. receivers there is no r -f by -pass Included across the plate supply rectifier for the set. If th e r e is an appreciable level of r -f signal on the power line feeding the re ce i v e r, r -f rectification in the power rectifier of the receiver can cause a particularly bad type of interference which may be received on other broadcast receivers in the vicinity in addition to the one causing the rectification. The soldering of a 0.01 -pfd. disc ceramic capacitor directly from anode to cathode of the power rectifier (whether it is of the vacuum -tube or selenium- rectifier type) usually will by -pass the r -f signal across the rectifier and thus eliminate the difficulty. Several sets have been encountered where there was only a slightly interfering signal; but, upon placing one's hand up to the volume control, the signal would greatly increase. Investigation revealed that the volume control was installed with its shaft insulated from ground. The control itself was connected to a critical part of a circuit, in many instances to the grid of a high -gain audio stage. The cure is to install a volume control with all the terminals insulated from the shaft, and then to ground the shaft. "Floating" Volume Control Shafts www.americanradiohistory.com 380 THE TV and Broadcast Interference r-i BO- RADIO METAL BAND 3.5 Mc. COIL, L CAPACITOR, 17 turns no. 14 enameled 100-;.,.íd. -inch diameter 23/4-inch length variable 3 11 7.0 Mc. C C TO A.C. LINE turns I no. 14 enameled 212 -inch 100- n,.fd. diameter length SHIELD BRAID LOV C - variable I Qv L TO TRANSMITTER OR RECEIVER I r SHIELD BRAID J l'2 -inch 14 21 and Mc. no. 4 turns 10 enameled 100 -,. ;.td. 3 -inch diameter 118 -inch length 3 27 and 28 Mc. turns I ¡ -inch o.d. copper tubing 2 -inch 1 variable diameter -inch length 100- ,,id. variable Figure 19 RESONANT POWER -LINE WAVE -TRAP CIRCUIT The resonant type of power -line filter is more effective than the more conventional "brute force type of line filter, but requires tuning to the operating frequency of the transmitter. Figure 18 COIL AND CAPACITOR TABLE FOR A -C LINE TRAPS a hen radio - frequency energy from a radio transmitter enters a broadcast receiver through the a -c power lines, it has either been fed back into the lighting system by the offending transmitter, or picked up from the air by over -head power lines. Underground lines are seldom rePower -Line Pickup sponsible for spreading this interference. To check the path whereby the interfering signals reach the line, it is only necessary to replace the transmitting antenna with a dummy antenna and adjust the transmitter for maximum output. If the interference then ceases, overhead lines have been picking up the energy. The trouble can be cleared up by installing a wave-trap or a commercial line filter in the power lines at the receiver. If the receiver is reasonably close to the transmitter, it is very doubtful that changing the direction of the transmitting antenna to right angles with the overhead lines will eliminate the trouble. If, on the contrary, the interference continues when the transmitter is connected to the dummy antenna, radio- frequency energy is being fed directly into the power line by the transmitter, and the station must be inspected to determine the cause. One of the following reasons for the trouble will usually be found: (1) the r -f stages are not sufficiently bypassed and /or choked, (2) the antenna coupling system is not performing efficiently, (3) the power transformers have no electrostatic shields; or, if shields are present, they are ungrounded, (4) power lines are running too close to an antenna or r -f circuits carrying high currents. If none of these causes apply, wave -traps must be installed in the power lines at the transmitter to remove r-f energy passing back into the lighting system. The wave -traps used in the power lines at transmitter or receiver must be capable of passing relatively high current. The coils are accordingly wound with heavy wire. Figure 18 lists the specifications for power line wave trap coils, while figure 19 illustrates the method of connecting these wave- traps. Observe that these traps are enclosed in a shield box of heavy iron or steel, well grounded. All -Wave Each complete- coverage home receiver is a potential source of annoyance to the transmitting amateur. The novice short -wave broadcast- listener who tunes in an amateur station often considers it an interfering signal, and complains Receivers accordingly. Neither selectivity nor image rejection in most of these sets is comparable to t ho s e properties in a communication receiver. The result is that an amateur signal will occupy too much dial space and appear at more than one point, giving rise to interference on adjacent channels and distant channels as well. If carrier- frequency harmonics are present in the amateur transmission, serious interference will result at the all -wave receiver. The harmonics may, if the carrier frequency has been so unfortunately chosen, fall directly upon a favorite short -wave broadcast station and arouse warranted objection. The amateur is apt to be blamed, too, for transmissions for which he is not responsible, so great is the public ignorance of short -wave allocations and signals. Owners of all -wave receivers have been quick to ascribe to amateur stations all signals they hear from tape machines and V- wheels, as well as stray tones and heterodyne www.americanradiohistory.com flutters. HANDBOOK The amateur cannot be held responsible when his carrier is deliberately tuned in on all -wave receiver. Neither is he accountable for the width of his signal on the receiver dial, or for the strength of image repeat points, if it can be proven that the receiver design does not afford good selectivity and image rean jection. If he so desires, the amateur (or the owner of the receiver) might sharpen up the received signal somewhat by shortening the receiving antenna. Set retailers often supply quite a sizeable antenna with all -wave receivers, but most of the time these sets perform almost as well with a few feet of inside antenna. The amateur is accountable for harmonics of his carrier frequency. Such emissions are unlawful in the first place, and he must take all steps necessary to their suppression. Practical suggestions for the elimination of harmonics have been given earlier in this chapter under Television Interference. Image Interference Interference Image In addition to those types of interference al read y discussed, there are two more which are common to superhet receivers. The prevalence of these types is of great concern to the amateur, although the responsibility for their existence more properly rests with the broadcast receiver. The mechanism whereby image production takes place may be explained in the following manner: when the first detector is set to the frequency of an incoming signal, the high -frequency oscillator is operating on another frequency which differs from the signal by the number of kilocycles of the intermediate frequency. Now, with the setting of these two stages undisturbed, there is another signal which will beat with the high- frequency oscillator to produce an i -f signal. This other signal is the so- called image, which is separated from the desired signal by twice the intermediate frequency. Thus, in a receiver with 175 -kc. i.f., tuned to 1000 kc.: the h -f oscillator is operating on 1175 kc., and a signal on 1350 kc. (1000 kc. plus 2 x 175 kc.) will beat with this 1175 kc. oscillator frequency to produce the 175 -kc. i -f signal. Similarly, when the same receiver is tuned to 1400 kc., an amateur signal on 1750 kc. can come through. If the image appears only a few cycles or kilocycles from a broadcast carrier, heterodyne interference will be present as well. Otherwise, it will be tuned in and out in the manner of a station operating in the broadcast band. Sharpness of tuning will be comparable to that of broadcast stations producing the same a -v -c voltage at the receiver. 381 The second variety of superhet interference is the result of harmonics of the receiver h -f oscillator beating with amateur carriers to produce the intermediate frequency of the receiver. The amateur transmitter will always be found to be on a frequency equal to some harmonic of the receiver h -f oscillator, plus or minus the intermediate frequency. As an example: when a broadcast superhet with 465 -kc. i.f. is tuned to 1000 kc., its high frequency oscillator operates on 1465 kc. The third harmonic of this oscillator frequency is 4395 kc., which will beat with an amateur signal on 3930 kc. to send a signal through the i -f amplifier. The 3930 kc. signal would be tuned in at the 1000 -kc. point on the dial. Some oscillator harmonics are so related to amateur frequencies that more than one point of interference will occur on the receiver dial. Thus, a 3500 -kc. signal may be tuned in at six points on the dial of a nearby broadcast superhet having 175 kc. i.f. and no r -f stage. Insofar as remedies for image and harmonic superhet interference are concerned, it is well to remember that if the amateur signal did not in the first place reach the input stage of the receiver, the annoyance would not have been created. It is therefore good policy to try to eliminate it by means of a wave-trap or low pass filter. Broadcast superhets are not always the acme of good shielding, however, and the amateur signal is apt to enter the circuit through channels other than the input circuit. If a wave-trap or filter will not cure the trouble, the only alternative will be to attempt OUTPUT INPUT T O CONSTANT K INPUT O O TYPE FREQUENCY OUTPUT T T M- DERIVED TYPE FREQUENCY Figure 20 TYPES OF LOW -PASS FILTERS Filters such as these may be used in the circuit between the antenna and the input of the receiver. www.americanradiohistory.com 382 TV and Broadcast Interference L, ANT L2 TO TC2 T` Cs RECEIVER ANT. POST TCA 1 O TO RECEivER CND. POST GND. Figure 21 COMPOSITE LOW-PASS FILTER CIRCUIT This filter is highly effective in reducing broadcast interference from all high frequency stations, and requires no tuning. Constants for 400 ohm terminal impedance and 1600 kc. cutoff are as follows: L,, 65 turns no. 22 d.c.c. closewound on 1%2 in. dia. form. L2, 41 turns ditto, not coupled to Lr. C,, 250 µµfd. fixed mica capacitor. C2, 400 Auld. fixed mica capacitor. C3 and C4, ISO Auld. fixed mica capacitors, former of S% tolerance. With some receivers, better results will be obtained with o 200 ohm carbon resistor inserted between the filter and antenna post on the receiver. With other receivers the effectiveness will be improved with a 600 ohm carbon resistor placed from the antenna post to the ground post on the receiver. The filter should be placed as close to the receiver terminals os possible. to select a transmitter frequency such that neither image nor harmonic interference will be set up on favorite stations in the susceptible receivers. The equation given earlier may be used to determine the proper frequencies. Pass Filters The greatest drawback of the wave -trap is the fact that it is a single- frequency device; i.e. -it may be set to reject at one time only one frequency (or, at best, an extremely narrow band of frequencies). Each time the frequency of the interfering transmitter is changed, every wave -trap tuned to it must be retuned. A much more satisfactory device is the wave filler which requires no tuning. One type, the low pass filter, passes all frequencies below one critical frequency, and eliminates all higher frequencies. It is this property that makes the device ideal for the task of removing amateur Low frequencies from broadcast receivers. A good low -pass filter designed for maximum attenuation around 1700 kc. will pass all broadcast carriers, but will reject signals originating in any amateur band. Naturally such a device should be installed only in standard broadcast receivers, never in all wave sets. Two types of low -pass filter sections are shown in figure 20. A composite arrangement comprising a section of each type is more effective than either type operating alone. A composite filter composed of one K- section and one shunt -derived M- section is shown in figure 21, and is highly recommended. The M- section is designed to have maximum attenuation at 1700 kc., and for that reason C, should be of the "close tolerance" variety. Likewise, C, should not be stuffed down inside L, in the interest of compactness, as this will alter the inductance of the coil appreciably, and likewise the resonant frequency. If a fixed 150 µµfd. mica capacitor of 5 per cent tolerance is not available for C1r a compression trimmer covering the range of 125175 µµEd. may be substituted and adjusted to give maximum attenuation at about 1700 kc. 19 -5 HI -FI Interference The rapid growth of high -fidelity sound systems the home has brought about many cases of interference from a nearby amateur transmitter. In most cases, the interference is caused by stray pickup of the r-f signal by the interconnecting leads of the hi -fi system and audio rectification in the.low level stages of the amplifier. The solution to this difficulty, in general, is to bypass and filter all speaker and power leads to the hi -fi amplifier and preamplifier. A combination of a VHF choke and 500 µpfd ceramic disc capacitors in each power and speaker lead will eliminate r -f pickup in the high level section of the amplifier. A filter such as shown in figure 17A placed in the input circuit of the first audio stage of the preamplifier will reduce the level of the r -f signal reaching the input circuit of the amplifier. To prevent loss of the higher audio frequencies it may be necessary to decrease the value of the grid bypass capacitor to 50 ppfd or so. Shielded leads should be employed between the amplifier and the turntable or f -m tuner. The shield should be. grounded at both ends of the line to the chassis of the equipment, and care should be taken to see that the line does not approach an electrical half -wavelength of the radio signal causing the interference. In some instances, shielding the power cable to the hi -fi equipment will aid in reducing interference. The framework of the phonograph turntable should be grounded to the chassis of the amplifier to reduce stray r -f pickup in the turntable equipment. in www.americanradiohistory.com 562 THE RADIO Receivers and Transceivers (figure 37). The front panel has the same dimensions as the outside of this box, and takes the form of a shallow pan, about 3/4 -inch deep (figure 36). The panel is affixed to two angle brackets mounted on the edges of the sub -panel. The two meters are mounted to the sub -panel, as is the dial mechanism and the pilot lamp. The various potentiometers are mounted to small L- shaped plates spaced away from the sub -panel. The pan -shaped panel is merely a decorative cover that finishes the appearance of the unit. The dial is home -made and is driven by a 35 -1 gear train made from re- mounted parts of a surplus BC -453 ( "Command ") receiver dial (figure 37). The dial drive and pointer may be made from a broadcast -type slide rule dial and the escutcheon is cut and formed from a piece of bakelite and is suitably engraved. Figure 38 UNDER -CHASSIS VIEW OF TRANSCEIVER Neat wiring and use of cabling techniques makes "clean" looking assembly. Power leads and long "runs" are laced into main cable passing in a square about r.f. section. Small components are soldered directly to socket pins, or are mounted on phenolic terminal boards, as is the case of the squelch components. R.I. coils and padding capacitors are at center of layout, beneath main tuning capacitor gang. Individual shield sections separate the r.f. stages. Change -over relay RY is mounted to rear wall of chassis next to antenna receptacle. www.americanradiohistory.com HANDBOOK Deluxe Mobile Transceiver 561 Figure 37 VIEW OF TRANSCEIVER WITH FRONT PANEL REMOVED The various panel controls are mounted on L- shaped brackets attached to the sub -panel by means of metal bushings. Meters are mounted to the sub -panel by means of encircling straps. The dial mechanism is made from geared portions of "command" receiver dial drive fixed to a thin phenolic plate. switches the 250 -volt supply from the receiver section to the transmitter section and section B transfers the antenna from the receiver to the transmitter. It is necessary to remove the B -plus from the modulator and power amplifier of the transmitter during reception, and this may be accomplished by switching off the high voltage supply by means of an auxiliary relay whose actuator coil is paralleled with the coil of relay RY1. The auxiliary relay should be located at the power supply. Transceiver Construction This transceiver is an excellent example of the fine workmanship possible by an amateur adept in sheet metal work and who has the necessary shop facilities. The chassis -cabinet is made of 14 -gauge sheet durai, cut and bent to size by a sheet metal shop. The assembly is made up of six pieces: A wrap- around back and side piece, removable top and bottom plates, the chassis, the sub -panel, and the front panel. Ventilation holes are drilled in the top plate and the wrap -around section to ventilate the unit, as a considerable amount of heat is generated by the tubes. The chassis is constructed with a 1/2 -inch lip around the edges which is bolted to the wrap- around piece and the sub -panel. In order to conserve height, the chassis has a "step" in it to allow room for the taller tubes (6146 and 6BQ6 -GT's) and the modulation transformer. Less room above the chassis is required for the receiver section, and a correspondingly greater area beneath the chassis allows room for the receiver coils and stage shields. The "step" can be seen in figure 36, running from the front to the back of the chassis, immediately to the right of the ganged tuning capacitors. The chassis, the wrap- around piece, and the sub -panel make up a complete TVI -proof box www.americanradiohistory.com 560 THE RADIO Receivers and Transceivers transformer coupled to two 6BQ6 -GT pentodes connected as zero bias class B modulators. No grid bias or screen voltage is required for the modulator, and the audio driving voltage is applied to the screens of the tubes. The control grid is connected to PLACEMENT OF MAJOR COMPONENTS the cathode. This simple circuit is capable of over 40 watts of audio output. Negative peak control is exercised by a silicon rectifier placed in series with the secondary winding of the modulation transformer, and a simple low pass audio filter composed of the leakage reactance of the modulation transformer plus the plate bypass capacitor of the r.f. amplifier stage reduces the higher order audio harmonics generated by this system. A high level of "talk power" is thus insured. The receiver section of the unit occupies the chassis, with the transmitter section at right. The "step" in the chassis is at the right of the main tuning Filament and Control Circuits Figure 36 TOP VIEW OF TRANSCEIVER SHOWS left -hand section of the gang, running parallel to it, from the front to the rear of the chassis. Receiver i.f. section runs along left edge of chassis, with squelch, regulator tube, and second conversion oscillator in the adjacent row. R.I. and audio stages are next to tuning gang. The three sections of capacitors nearest the front panel are for the receiver portion, while the rear capacitors are for the transmitter section. The 6146 plate tank coil is at the rear of the chassis. Modulator section occupies right -hand portion of chassis, with transmitter r.f. stages immediately to the left. 6CL6 buffer tube is shielded and directly in front of the 6146. The transmitter is designed for either 6- or 12 -volt operation. The circuit of figures 34 -35 shows 6 -volt configuration. For 12 -volt operation, it is only necessary to rewire the power plugs as shown and filament switching is automatic. Change -over from receive to transmit is accomplished by means of relay RY1, which is actuated by the microphone button, and which has a d.c. coil suited to the voltage of the automobile battery. Contact section A of this relay www.americanradiohistory.com zo d: rß F' ú W ^ °Éo ^ C °O °^ Y ° N - m ai é á i ñ °O s ^ó °a° ... . Ó P1 « PI ro v ° ó ° p .ú E u °I h. ë aMó'°e e o e . °^ éa` tè° ú M U Ó ' Ñ 4 # M ó Ñ E o N° TLL R J ;!1,f, N I OY 11 -- o- ÓYN Y 8Y .71(-111 8 8Y YIs N Hu T 2 www.americanradiohistory.com ry IHI' =3 h `- h N J ` IL R ° F . , g v m ú ° ó ".Y N NY Hi' 3-; . .9. Z.- 1 28 òó 0 > ` O ù t ....Yu E z,- ° chÉe E -ó« - e b =â =o 3 ó ldd o°o CO 11 *f If 7 0; 1 O..~. ó c ö o á . .E _ « ó E o = E ; n «.,>. ^ a a ° Y : = o.a Gn,°.x¿^r t E.:.x ° . U _ _ v . U E 3 o N O rO O J M 7 GA lo°oocJ ò LL y . i'o %O ti M ac ì, ; tc r ` U U E É c n-I ,9 '°o Q..00(2, N U > f ¢ ;e (00000 _ V in W Ñ Yo vW+- 11 II--II u J www.americanradiohistory.com ° i `` r O 't .a w V Ó O óh ^aJ 7 0 ó C m-E a3, U U C- 7 E E aÉ fic C :jc^ ó°`E \_;mú á = 3 r cU 0 .. P 4E ] nf 0 rn '0000 ti W c C o Q OZ n c 0 n W t- c m O V I^ ° 7 -3 W2 CO ú ú p a 'f° c 1 0L-Ti t., ry .ro' 2a °`t 0 M U ~ . : tnW i] V I W 2 IE-tf V Z u O O I~IFI ;I .^ hr ax N 2 ,; =E=Er ' ' ó O v ó 4Ó O U EañçúEoúoEE =1sIIiu J J J HANDBOOK Deluxe Mobile Transceiver stability. The oscillator runs continuously and is voltage regulated. A single i.f. transformer (T1) provides sufficient image selectivity at 4.26 Mc. and no additional amplification or tuned circuits are required. The second intermediate frequency is 260 kc., and a 6BE6 multi -grid converter tube is used as a mixer to this frequency. The local oscillator is crystal controlled at 4.52 Mc., and makes use of the 6BE6 as a "hot cathode' crystal oscillator. Precise adjustment of the oscillator frequency may be made by means of the variable inductance (L;) in the grid- cathode circuit of the mixer tube. The choice of frequency of the mixing oscillator is important in that no harmonic frequencies of the oscillator should fall into the 10 meter band, or into its "image" frequency band. This insures that undesired "birdies" or spurious responses of the receiver are reduced to an absolute minimum. Two stages of i.f. amplification employing low filament drain 6BJ6 tubes provide sufficient receiver gain, and are followed by a 6AL5 detector/a.v.c. rectifier stage. A two -stage noise limiter patterned after the popular "twin noise squelch (TNS) circuit" provides maximum noise rejection with minimum audio distortion. A 12AX7 and 6AL5 are used in this portion of the receiver. A 12AT7 tube serves a dual purpose as a first audio stage and v.t.v.m. -type S -meter amplifier, followed by a 6ÁQ5 audio output stage. The S -meter circuit makes use of a "backwards reading" meter that rests at full scale. The a.v.c. voltage applied to the amplifier tube reduces the meter current in accordance with the strength of the incoming signal. The Transmitter Section. The transmitter section of the transceiver is shown in figure 35, and in outline form in figure 32. A 12AT7 dual triode serves as a mixer -oscillator stage, beating the receiver v.f.o. with a 4.26 Mc. crystal (equal to the receiver intermediate frequency) The sum of these two frequencies is the transmitting frequency, which is equal to the frequency of reception. Following the mixer -oscillator are two gang -tuned r.f. amplifier stages employing high gain 6CL6 pentode tubes. The second stage is neutralized for maximum stability. The power amplifier stage uses a single 6146 in a pi- network output circuit, which is also gang -tuned in conjunction with the exciter and v.f.o. Tuning and loading controls of the power amplifier stage are located on the rear of the chassis and need not be readjusted unless a change is made in the antenna system (figure 39) Antenna change -over is controlled by a section of relay RYI. Grid and plate currents of the 6146 are monitored by meter M2. The Modulator Section. The modulator is designed to work with either a ceramic -type crystal microphone, or a high impedance dynamic unit. A 12AT7 serves as a two stage resistance coupled amplifier, exciting a parallel connected 12ÁU7 driver. This, in turn, is . . Figure 33 MINIATURE POWERHOUSE PACKS PLENTY OF PUNCH! The transceiver is in a custom -made 557 built case which permits maximum utilization of available space. Mounting flanges may be seen attached to upper portion of transceiver case. At left of main tuning dial are volume control (with on -off switch), r.f. gain control, and squelch. At right are microphone level control, meter switch, and microphone receptacle. riwgr- www.americanradiohistory.com THE RADIO Receivers and Transceivers 556 ANT R.F. AMP. 1ST MIX (26 Mc.) (4.26 AK.) ® ® 2NDI%. (280 I.F. I.F DET. AVC RC. ® ® ® ® GAIN AUDIO AUDIO 5-METER 12AT7 ®'I SPMR (4.26MC) SPEECH AMP. DRIVER J MIC Figure 32 BLOCK DIAGRAM OF THE TRANSCEIVER The tuning oscillator of the unit covers the range of 23.74 -25.44 megacycles. The transmitter conversion crystal (4.26 Mc.) is the same frequency as the first i.f. of the receiver, thus placing receiver and transmitter operating frequencies at the same soot on the tuning dial. Receiver selectivity is obtained by use of two i.f. stages at 260 kc. R.I. circuits of both transceiver sections are ganged for single dial control. Circuit A block diagram of the trans - ceiver is shown in figure 32. The circuit utilizes a double conversion receiver employing eleven tubes and a voltage regulator, and a v.f.o.- controlled amplitude modulated transmitter having eight tubes. A feature of the unit is that transmitting and receiving frequencies are locked together and controlled by one master oscillator. All variable r.f. circuits are tracked for single control tuning. The operator merely tunes the transceiver to the station he desires to contact, pushes the microphone control button and the transmitter is tuned to the same frequency, ready to "talk." The Receiver Section. The receiver portion of the transceiver is shown in figure 34, and in outline form in figure 32. Double conversion is used, with the second conversion oscillator crystal controlled. The first conversion oscillator is also the v.f.o. for the transmitter section, as explained later. The three r.f. circuits Description of the receiver section (r.f. stage, mixer, and oscillator) are gang -tuned for proper tracking across the 10 meter band. The r.f. stage utilizes a 6BZ6 high gain, semi- remote cutoff pentode to achieve maximum signal gain without troublesome cross modulation effects from strong nearby signals. The circuit of this stage is conventional, except that the cathode return may be removed from the gain buss by switch S) for optimum weak signal response, if desired. Partial a.v.c. is applied to the 6BZ6 by means of a high impedance voltage divider in the a.v.c. system. A 6BA7 multi -grid converter tube is used as a mixer from the operating frequency to the first intermediate frequency of 4.26 Mc. Mixer injection voltage is applied to the #1 grid of the 6BA7. The local oscillator employs a 6ÁH6 and tunes the range of 23,74025,400 kc., with a slight overlap at both ends of the range. A high -C "hot cathode" oscillator circuit is employed for maximum frequency www.americanradiohistory.com HANDBOOK Deluxe Mobile Transceiver shown in figure 30. This unit is sufficient to run one converter at a time. 27 -6 560 10N 8+ 70 V. B+RCG A Deluxe Mobile Transceiver The modern automobile leaves little room for radio equipment mounted in proximity to the driver. Mobile equipment, as a result, must be built more compactly in order to fit in the dashboard firewall area available for auxiliary equipment. The amateur having sheet 555 6.3 Ti = V. 125 V., 50 MA. 6.3v.,2A STANCOR PI-6121 SR= SELENIUM RECTI FIER. S0 MA. GND Figure 30 SCHEMATIC, CONVERTER POWER SUPPLY Figure 31 COMPACT TRANSCEIVER OFFERS ULTIMATE IN MOBILE COMMUNICATION This compact a.m. transceiver is a complete meter station, packaged so that it will fit into all but the most cramped automobiles. The transmitter section runs up to 70 watts input and is designed for "on frequency" operation with the receiver section. The easy to -read dial controls the master oscillator for both transmission and reception. The operator merely tunes the transceiver to the station he desires to contact and the transmitter is automatically tuned to the correct frequency. The transceiver is mounted in the car by means of dashboard clamps fastened to the top of the unit by means of a sliding fixture. Top and bottom plates are removable by means of snap fasteners, and are perforated for good ventilation. Simplicity of operation permits transceiver to be operated without the driver taking his eyes from the road. 10 metal working facilities at hand is indeed fortunate, as he may custom-form his equipment chassis and cabinet to fit the space provided in his particular automobile. Described in this section is a deluxe transceiver, designed and built by W7JNC which will fit easily into all but the most cramped automobiles. The unit is a complete 10 meter station capable of running up to 70 watts input, having a sensitive double conversion receiver, and packaged in a cabinet measuring only 11 inches wide, 4 inches high, and 8 inches deep. The transceiver is suited for either mobile or fixed -station operation. www.americanradiohistory.com 554 Receivers and Transceivers THE RADIO Figure 28 PLACEMENT OF MAJOR COMPONENTS ABOVE THE CHASSIS Figure 29 CLOSE-UP OF 2 -METER R.F. AMPLIFIER STAGE A shield partition passes across the center of Nuvistor socket. The grid compartment is at the right, and plate cornpartment at the left. Coil L, is wound on high value composition resistor. Six meter r.f. section is identical except for coil changes. I, 1 www.americanradiohistory.com www.americanradiohistory.com 552 Receivers and Transceivers verter. Adjust the converter output coil (L6 /L16) for maximum receiver noise, making sure that you are not tuning to the image frequency of the receiver. Connection to the receiver should be made by means of a short length of coaxial line to prevent spurious signal pick -up in the 28 -30 Mc. range. With these preliminary adjustments made, the r.f. stage is ready for test and alignment. Start with the 144 Mc. section. Remove the B+ to coil L3 and insert a 6CW4 in the r.f. socket. Connect a temporary antenna to the converter and tune in a strong local test signal. Make sure signal pickup is via the antenna and not by indirect pickup via coils L3 or L5. Roughly peak coils L3, L ;, and L6 for Figure 26 UNDER -CHASSIS VIEW OF "SIAMESE" CONVERTER The converter chassis has been removed from the end plates for this photograph. The two crystal oscillators are at the center of the chassis, with the mixer stages adjacent to them. At the ends of the chassis are the r.f. amplifiers. Note that a T- shaped shield iso- lates the input and output circuits of the r.f. amplifier from the remainder of the circuitry. The shields are made up of thin flashing copper and are about 112 inches high. The small leg of the shield passes across the center of the Nuvistor socket, and the grid -plate blocking capacitor passes through a hole drilled in this partition. maximum signal. Now, carefully spread and adjust the turns of coil L2 for minimum received signal. The neutralization point will be a sharp and almost complete signal null. If neutralization is obscure, add or remove a turn or two of wire from coil L2. Now, reconnect the B -plus lead to the plate coil of the r.f. stage and tune in a weak signal near the center of the desired tuning range. Peak coils L3, L5, and L6. Coil L1 will tune very broadly. Recheck the neutralization once again (after removing the r.f. B -plus lead) and secure the turns of coil L2 with a spot of cellulose cement or colorless nail polish. As a final check, measure the plate current of the r.f. stage. It should run approximately 8 ma. and should not vary when the antenna is disconnected from the stage. A variation in plate current indicates oscillation of the r.f. amplifier. If a noise generator is available, coil L1 and the antenna tap can be adjusted for a onedecibel or so improvement in noise figure after the above adjustments are completed. Adjustment of the six -meter converter is identical to the above outline. Plate power requirements of each converter are 70 volts at 8 ma. for the r.f. stage, and 105 volts at approximately 10 ma. for the mixer and oscillator. A suitable supply is The Converter Power Supply www.americanradiohistory.com A -I- + o # + I -Fink Ì + -Iv + +v I H- + _IN + + v www.americanradiohistory.com Receivers and Transceivers 550 Figure 24 SCHEMATIC, "SIAMESE" CONVERTER FOR 2- AND 6- METERS -5 -512 turns hookup wire close wound about Id. trimmer capacitor. the body of 6 L9 -19/s turns :32 e. close wound about the trimmer capacitor. 6 ,o.fd. of body L,0-16 turns :28 e., tap 51/2 turns from ground. Some as 1.1. L,1- Neutralizing coil. 50 turns :36 e., wound turns :26 e., on 14 -inch Coil data: LI, L3 diameter polystyrene rod or form. Wind 3/8 -inch long, top 2 turns from ground end on L,. Adjust by spreading turns. L2-Neutralizing coil. 20 turns :30 e. on 10 megohm resistor, close 5 :32" diameter wound. Adjust by spreading turns. L4 turn hookup wire over 8 -plus end of coil L3. L5-6 turns, some as L,. Tap 2 turns from ground end. L6, 116 -26 turns :32 e. on 1/4-inch slug -tuned form, close wound. (Cambridge :PLS -6 v.h.f. form with green colored slug.) turns hookup wire over 8 -plus end of L7, LIT Coils L6 and L,6. L8 -I some as L2. L,2 -19 turns :28 e., some construction os L6. turn hookup wire over B -plus end of L,3 coil L12. L14 -Some as L,3, wound over ground end of coil L15. L,5 -17 turns :28, same construction as L6. L18-25 turns :32 e., same construction as L6. L19 turns hookup wire over 8 -plus end of coil L16. See text. -1 -2 -2 vl 6Cwa vz 6AK5 A. F L2 C2,500 MIA L3 LI L6 LS O 4 /44-146 MC °411. Ca, L 1= C3 MC. C6 C 20 000 500 1000 500 2B-30 9 V3 6AK5 Yac X1 38.66 MC c C7 1000 P, Br 70V. % t I V4 6Cw4 Ln O Q p = C23 500 GND. I 6.3V. 2 Sz 500 MIA. Lis Lis 5 Cu L13 Lu ° 50 -52 MC. RCS. 51 50 MCI 6AK5 Li2 Lio MC.I Vs A.F. C12,500 8+105V ` --o A- IMF -T- -3 500 2 8-30 MC. Re 2206 Cis Cis 000 1000 NOTES Ve I. ALL RESISTORS i /2-WATT. 6AK5 2.500 ULF CAPACITORS ERIE OSC. GP -500 SILVCR MICA BUTTON. I1 3. 1000 LUF CAPACITORS, ERIE GP -500 SILVER MICA BUTTON. 4. 61.UF TRIMMER CAPACITORS CENrRALAB B2P -6 OR EOUIV. CINCH-JONES l33-65 -10 -001. 5. NUVISTOR SOCKETS 6. FILAMENT BYPASS CAPACITORS: CENTRALAB DISC L19 X2 22.0 MC. 0 HIR; 10501 C25 500 Cie 1000 www.americanradiohistory.com 100K= Re, 470 HANDBOOK 2 -6 M. "Siamese" Converter of the triode -connected 6AK5 operating with grid injection. The link circuit from the r.f. amplifier stage is tapped directly to the 144 Mc. mixer coil to obtain optimum coupling. The Local Oscillator Stage. A 6AK5 tube is used as the crystal controlled oscillator in the 2 -meter converter. A 38.66 Mc. overtone crystal oscillates in a grid- screen circuit, with the plate circuit tuned to the third harmonic (116 Mc.). The oscillator is capacitively coupled to the grid circuit of the mixer stage. The six -meter converter makes use of a 6AK5 overtone oscillator using a 22 Mc. crystal. The tube is connected as a triode, and the oscillator is inductively coupled to the cathode circuit of the mixer stage. This configuration is required to obtain sufficient injection voltage without permitting the 22 Mc. frequency to appear in the broadly tuned plate circuit of the mixer. While the pentode mixer is undoubtedly noisier than the triode, the overall noise figure of the converter is much less than the atmospheric noise at 50 Mc. so this configuration does not tend to degrade the usable sensitivity of the converter. 549 All chassis holes are drilled, the major components mounted in place, and then the auxiliary shields are soldered to the chassis. Placement of parts may be seen in figures 26 -29. The six tube sockets lie along the center line of the chassis and all wiring is done in a point to -point fashion. The 500 -µpfd. ceramic grid plate blocking capacitors pass through small holes drilled in the interstage partitions and are supported between the top terminal of the r.f. stage plate coil and one lead of neutralizing coil L_ The neutralizing coil, in turn, is attached to the top (grid) terminal of the r.f. stage grid coil. Every effort should be made to make all leads in the r.f. stages as short and direct as possible. Mixer stage wiring is straightforward. The cathode injection coil of the 50 Mc. mixer may be made of a length of small hook -up wire run from pin #2 of the 6AK5 socket, looping twice around oscillator coil L1R, then back to the 6AK5 socket, to be soldered to the grounded filament terminal of the socket. Lii. Testing the Wiring should be checked and Converters Converter Construction As each converter is extremely small in size, it is simple to construct both of them upon a single chassis. The two units are therefore mounted on a small copper plate measuring 3" x 7" in area, having a 1/2-inch turned down lip running along the edges. A drilling template for the chassis is shown in figure 25. If the sheet copper is not available, a phenolic "printed circuit board" covered with a thin layer of copper may be used as a substitute. the mixer and oscilator tubes placed in their sockets. Power is applied to the converter and the oscillator stage adjusted for operation. A grid -dip oscillator or a nearby receiver will serve as a handy indicator of oscillation. You can temporarily unground the 220K grid resistor of each mixer stage and insert a low range micro- ammeter in the circuit, tuning the oscillator controls for maximum mixer grid current. If a v.t.m. is handy, it may be attached to the grid pin of the mixer stage and the oscillator controls adjusted for maximum negative grid voltage. Voltage should measure between and -1 Figure 23 THE "SIAMESE" CONVERTER PROVIDES SUPERIOR V.H.F. PERFORMANCE ON TWO BANDS volts. -2 Next, connect a receiver capable of tuning the 28 -30 Mc. range to the output of the con- This dual converter has a noise figure better than 3 decibels on 2- and 6- meters. Utilizing crystal control for maximum frequency stability and the new Nuvistor triode, superior performance is achieved at minimugp cost. The converter is built upon a small copper chassis mounted to an aluminum panel by means of two end plates. Panel size is 31/4" x 81/2 ". Plate voltage is applied to both converters, and filament voltage is controlled by the panel mounted toggle switches. Nuvistor tube (right) is compared to conventional 6AK5 in foreground. www.americanradiohistory.com 548 THE RADIO Receivers and Transceivers Figure 21 EASE OF MOUNTING IN YOUR AUTOMOBILE IS FEATURED IN THESE UNITS The transceiver and supply have low that they may be placed in line beneath power profile so the dashboard of your automobile. Tuning controls are easily accessible to driver of car. of these converters is better than 3.5 decibels, which compares favorably with units employing the expensive 417A low noise triode, and may only be surpassed by use of the costly 416B tube. For simplicity and ease of operation, the two miniature converters are built on one panel- chassis combination approximately 8" x 31/2" in size. The units may be powered from the communications receiver, or may be run from a separate supply as desired. The circuits of the two con Description verters are similar except for minor details ( figure 24). A 6CW4 is used as a grid driven, neutralized r.f. stage, link coupled to a 6AK5 mixer stage. A second 6AK5 serves as a crystal -controlled local oscillator. The intermediate frequency range is 28 to 30 Mcs. The choice of a high i.f. eliminates image problems and permits use of a simple slug -tuned coupling circuit between the converter and the companion Circuit The Mixer Stage. A pentode- connected 6AK5 serves as a grid biased mixer for the 50 Mc. converter. Cathode injection from the crystal controlled local oscillator is used to achieve proper mixing voltage. Use of a triode mixer stage is not recommended as the reduction in conversion noise of the triode over the pentode is minimal at 50 Mc. and there is tendency of the triode to regenerate as the frequency of the injection oscillator is quite close to the intermediate frequency. The 144 Mc. mixer stage takes advantage of the lower mixer noise level Figure 22 THE RCA "NUVISTOR" VHF TUBE The miniature RCA Nuvistor triode provides high gain, low noise performance in the v.h.f. spectrum at low cost. Intended for TV use, this small tube shows excellent results in the 2- and 6 -meter converter described in this section. receiver. The R.F. Stage. The r.f. stage of each converter consists of a single 6CW4 Nuvistor triode. Inductive neutralization is used (L2 and L11) incorporating a series blocking capacitor to remove plate voltage from the circuit. This simple configuration provides above 20 decibels of usable gain, which is more than sufficient to override mixer noise. A single stage such as this is noticeably less susceptible to cross -modulation from strong local signals than is a double stage (6BQ7A, for example) , or two cascaded high gain stages. www.americanradiohistory.com HANDBOOK 2 -6 M. "Siamese" Converter milliameter across the plate "test" points. A 20 watt lamp bulb may be attached to the antenna receptacle as a dummy load. Power is now applied and the pi- network circuit is adjusted for maximum glow of the lamp. A 0 -10 d.c. milliameter placed across the grid "test" points may be used to adjust the excitation level to the 2E26. Grid current should run between 2 and 3 ma., and plate current is approximately 50 ma. For 21 Mc. operation, the "grid tuning" capacitor is resonated to 21 Mc. and the pinetwork retuned to this band. Slight adjustment of L3 and L4 will permit the two bands to be properly tuned by swinging the resonating capacitors from minimum to maximum capacitance. The last step is to switch to v.f.o. operation, and adjust the slug of coil L1 for proper dial calibration. The slug should be permanently fixed in position with a drop of nail polish to prevent mechanical instability during mobile operation of the unit. The "magic eye" tube can be used to indicate amplifier resonance, but an external plate meter is recommended for v.f.o. operation, since loading must be readjusted as the transa 0 -100 d.c. 547 mitter frequency is varied. The "eye" tube can be used for loading adjustment, but it takes practice to interpret variations in the pattern. The Power Supply An inexpensive power supply suitable for a.c. operation is shown in figure 20. Voltage regulation is employed for maximum stability. Mobile supplies, such as the transistor types shown in the Power Supply chapter are suitable for mobile operation. Low voltage required for operation of the v.f.o. and receiver may be obtained from a dropping resistor and regulator tube. 27 -5 "Siamese" Converter for Six and Two Meters The new R.C.A. Nuvistor series of miniature tubes brings low noise level v.h.f. reception within the economic capability of the average radio amateur. Described in this section are twin crystal controlled converters for 50 and 144 Mc. that make use of the 6CW4 Nuvistor v.h.f. triode. The inherent noise level Figure 20 HOME MADE POWER SUPPLY FOR TRANSCEIVER FITS IN MATCHING CABINET WITH SPEAKER Simple transformer-oper ated a.c. supply is used for home station work. VR -ISO provides regu- lated voltage for maximum stability. Dynamic speaker is included in enclosure. www.americanradiohistory.com 546 THE RADIO Receivers and Transceivers speaker to the audio jack. Light the filaments and apply plate voltage. Transformers T1, T2, and T3 can be aligned by loosely coupling a 2050 kc. signal from an external source to the plate circuit (pin #6) of the 6CG8 mixer tube. Next, the bandswitch is placed in the 10 meter position and a 28 Mc. signal is applied to the input circuit of the receiver. Proper tracking is achieved in the usual manner, with the oscillator padding capacitor determining the calibration at the high frequency end of the dial, and the variable slug of the oscillator coil (L6) being used to set the edge of the band at the low frequency end of the dial. An Figure 19 REAR VIEW OF UNDER -CHASSIS AREA Point to point wiring is used, with many small components soldered directly to the tube socket pins. Coaxial antenna receptacle, microphone receptacle, and crystal- v.f.o. switch are mounted on back apron of chassis. Pilot lamp receptacles are bolted to frame of tuning capacitor which is dropped below the chassis deck by means of cut -out in deck. antenna can now be connected to the antenna jack of the transceiver and signals should be heard. Mixer plate coil L, is peaked for maximum signal response near the center of the band and adjustment of the transmitter pinetwork circuit can be made for greatest receiver sensitivity. Bandswitch S2 is now placed in the 15 meter position and the oscillator padding capacitor is adjusted to correctly position the high frequency end of the 15 meter band when the tuning capacitor is at minimum setting. The mixer padding capacitor is adjusted for maximum receiver sensitivity at the same frequency. The transmitter portion should now be aligned. Place the 6AU6 and 6CL6 tubes in their respective sockets and insert a 7 Mc. crystal in socket X2. Throw S1 to the transmit position and adjust Ln for proper crystal oscillation. Next, plate coil L3 of the 6CL6 stage is adjusted to 28 Mc. with the "grid tuning" capacitor nearly open. Plate voltage is removed and the 2E26 is inserted in its socket. Place www.americanradiohistory.com HANDBOOK Meter Transceiver 10 -15 of the stator support bars leaving them attached to one bar. Cut the rear plate so that it is supported only by the other bar. A small planetary unit is placed between the capacitor and the dial for ease of tuning. A second planetary unit is used for the transmitter v.f.o. All sockets, terminal strips, and trimmer capacitors are mounted in place using 4 -40 hardware with soldering lugs placed .,eneath the nuts in various convenient positions. Transceiver Wiring The wiring of the unit is quite simple if done in the proper sequence. The under -chassis area contains many small components but these need not be crowded, provided proper care is taken in the layout and installation of parts. The smaller components (capacitors and resistors) are installed between the socket pins of the various tubes. Socket ground connections are made before the wiring is done, filament wiring is done next, then the socket -mounted components are placed in position. Number 22 stranded thermoplastic insulated wire (0.07" diameter, Consolidated #737) is recommended for all leads except the filament circuit. Number 18 wire should be used for these leads. Small diameter, insulated phono- type" shielded wire is used for the lead running from pin #2 of the 12AX7 to the receiver volume control capacitor. The filament circuit is wired in a series -parallel arrangement so that either 6- or 12 -volt operation may be chosen at the power plug. Before i.f. transformer T1 is mounted in position, it should be modified so that it tunes to 2050 kc. Some makes of transformers will reach that frequency with no modification. Others will require that some turns be removed from the primary and secondary windings, or that the value of internal fixed capacitance be reduced accordingly. The three small 15 meter variable padding capacitors are mounted below the chassis in close proximity to the bandswitch and may be seen in the center of the chassis (figure 17) Crystal socket X1 is mounted horizontally on a small metal bracket under the rear of the chassis so that the type FT -243 crystal may be inserted and removed from the rear of the transceiver. The buffer tuning capacitor (marked "grid tuning" on the front panel) is mounted on a small aluminum bracket at the middle of the chassis. Oscillator coil L1 is posi- ' . - 545 REAR PANEL REAR EDGE Or CHASSIS Ta xl TUNING T1 T -R CAPACITOR I RDC6 SWITCH 6E5-M I PLATE !TUNING/ 1 DIRONT ii 5UBPANEL L FRONT U LIRONT PANEL EDGE Or CABINET Figure 18 LAYOUT OF MAJOR COMPONENTS ABOVE THE CHASSIS tioned between the buffer capacitor and the v.f.o. capacitor, and is mounted in a small aluminum shield cut down from an i.f. transformer can. A dust plate is bolted to the rear lip of the chassis and adds extra strength to the assembly by virtue of the two angle brackets bolted to the plate and chassis. The 2E26 socket is mounted on this plate, as are the v.f.o., crystal switch, power plug, and antenna receptacle. A slot is cut along the top edge of the dust plate to insure adequate ventilation. Transceiver Coils Only six coils are required for the transceiver, four of them in the transmitter section. Because of the compact construction and the influence of nearby objects, it is wise to grid -dip each coil to the proper frequency after installation. Oscillator plate coil (L2) is resonated by the internal capacitance of the associated tubes and stray circuit capacitance, and should be grid dipped with the oscillator and buffer tubes in their respective sockets. Testing the Transceiver The transceiver should be tested a section at a time. Start with the receiver and audio portion. Insert the tubes in the sockets and place crystal X1 in the holder and connect a temporary www.americanradiohistory.com 544 THE RADIO Receivers and Transceivers tank circuit is very short. Directly behind the tuning capacitor is the modulation transformer. The 2E26 transmitting amplifier tube is mounted in a horizontal position at the right of the chassis, as shown in figure 15. The transmitter pi- network output circuit is panel mounted, directly in front of the plate cap of the 2E26. The 6ME -10 tuning "eye" is panel Figure 17 UNDERCHASSIS VIEW OF TRANSCEIVER V.f.o. tuning capacitor and coil (in shield) are at top edge of chassis. "Grid" tuning capacitor is recessed behind panel and driven with shaft extension. The 21 Mc. padding capacitors are directly behind bandswitch at center of chassis. 1.1. amplifier is along chassis edge in foreground. mounted between the pi- network components and the receiver tuning capacitor. Placement of receiver components is conventional, with the 6CG8 mixer stage mounted near the tuning capacitor. The receiver tuning capacitor is a two section unit, converted from a single section Johnson 167 -3 variable capacitor. Using a small hacksaw or jeweler's saw, the two stator rods are cut so that a front group of plates are supported by one post, and a rear plate is held by the other post. This is the way you do this operation: Leave the front two stator plates and the rear stator plate in position, removing the three other plates in between. Next, cut the remaining two front plates away from one www.americanradiohistory.com 3 www.americanradiohistory.com 542 Receivers and Transceivers section. During the transmission the "eye" indicates proper amplifier adjustment. The three stage audio amplifier serves as a modulator for the transmitter as well as an audio system for the receiver. During transmission, both sections of a 12AX7 serve as a voltage amplifier, driving two 6CM6 pentode tubes in a parallel class A modulator circuit. A simple resistive feedback circuit from the plates of the modulator to the plate of the driver stage improves speech quality and reduces distortion. The audio output transformer T4 serves as a modulation choke when switch section S1.1) opens the return circuit of the loud speaker jack. Switch section SI_c couples the modulator to the plate circuit of the r.f. amplifier stage. In the receiving mode, the audio signal from the diode second detector circuit is applied through the volume control to the grid circuit of the second section of the 12AX7 speech amplifier. The cathode circuit of the first section of the 12AX7 is opened by switch section S1.1) during reception. Transceiver Layout Figures 13, 15, 17 and 19 illustrate the general plan of the transceiver. The panel layout of controls is shown in figure 13, and parts placement above the chassis is illustrated in figures 15 and 19. The transceiver is built upon an aluminum chassis 67/8" x 55/8" x 1" in size. This assembly fits within a steel wrap- around type cabinet .33,4" high, 63/8" deep and 7" wide. This cabinet was custom -made to allow absolute minimum size of the transceiver. A manufactured cabinet can be used at a sacrifice in compactness. The California Chassis Co. type LTC-464 cabinet and chassis, with an over -all measurement of 41/2" x 91/8" x 71/8" is suitable and less expensive than a custom package. and Assembly The transceiver makes use of a dual front panel. Both panels are made of 1/16 inch clear plastic sheet. The sub -panel is bolted directly to the chassis and is painted black to provide a good background for the tuning dial. The front panel is a similar piece of plastic, spaced about 1/4 inch in front of the sub -panel by means of four bolts and metal spacers. This panel is painted and lettered as shown. For decorative purposes, a thin strip of aluminum is run across the bottom of the panel to provide a pleasing color contrast to the eye. Layout of principal parts above the chassis can be observed by comparing the photographs with figure 18. Viewed from the top front, the receiver occupies the left portion of the chassis and the transmitter occupies the right half. The external plugs and receptacles are mounted on the rear apron of the chassis. The receiver tuning capacitor is centered on the chassis, with the 6DC6 r.f. amplifier tube mounted horizontally above it on a bracket. The socket is oriented so that the grid connection between the tube and the amplifier Figure 16 SCHEMATIC OF TRANSCEIVER Receiver tuning capacitor-Oscillator section 3 -18 ppId. Detector section 3 -8 µµId. (See text for details.) Pi- network loading capacitor -400 µµId. Allied Radio Co., Chicago, 111. #61 -H -009. L, 1H. 1/2" diam. form, 1" long, tuned with adjustable 1/4-20 iron core slug. (7.07.42 Mc.) Wind with # 18 e. L2 -25 µh. Tunes to 7 Mc. with circuit capacitance. 5/16" diam., 1/2" long with adjustable 1/4 -20 iron core slug. Wind with -1 22 e. L3-0.9 pH. Tunes to 21 Mc. with tuning capacitor at maximum, and 29.7 Mc. with capacitor at minimum. 5/16" diam., 1/=" long with adjustable 1/4 -20 iron core slug. Wind with »22 e. L4-0.9 µh. Tunes to 21 Mc. with tuning capacitor at maximum, and 29.7 Mc. with capacitor at minimum. B&W coil, 3/4" diam., 8 turns per inch #18 wire. L5-1.5 µH. Tunes to 21 Mc. with tuning capacitor at maximum and auxiliary padding capacitor in circuit, and 29.7 Mc. with tuning capacitor at minimum and auxiliary capacitor out of circuit. -Two windings. Tuned winding: 0.4 pH, wound on 5/16" diam. form., 1/2" long with adjustable 1/4 -20 iron core slug. Wind with .-.18 e. Secondary winding: 0.4 pH scramble wound, spaced 1/8 -inch from tuned winding, #22 d.c.c. Tunes 25.95 -27.65 Mc. for 10 meters, 22.05 -22.5 Mc. for 15 meters. Note: Coils may be wound on J. W. Miller Co. #41 -A000 -CBI ceramic forms, with type R slug. Alternatively, J. W. Miller Co. #20A and #21A series adjustable r.f. coils may be substituted. All coils should be adjusted to frequency with o grid -dip oscillator. T1 -2050 kc. i.?. transformer. J. W. Miller Co. #13-WI. Remove turns from 1500 kc. windings to resonate at 2050 kc. T2T3 -265 kc. i.f. transformer. J .W. Miller L6 Co. #12-H1. 74- Primary, 10 -watt. 5000 ohms. Secondary 4 ohms. Dial -Made up of Jackson Bros. planetary drive. (Arrow Electronics Co., 6S Cortland St., New York 7, N.Y.) Tuning Eye: 6ME -10 (midget) or EM -84. (See tube manual for pin connections.) www.americanradiohistory.com HANDBOOK 10 -15 semi - remote cutoff pentode is used as an r.f. amplifier with the input grid connected directly to the r.f. circuit of the transmitter power amplifier. Thus, when the transmitter is properly adjusted and loaded to the antenna system, the receiver input circuit is automatically tuned to the same frequency. This eliminates the components and space normally required for a tuned r.f. input circuit. The coupling capacitor and grid resistor of the 6DC6 stage are chosen so that the tube blocks itself off during transmission periods. The relatively large potential developed on the grid of the r.f. stage does no harm. A 27 -ohm composition resistor is placed in the plate lead of the r.f. amplifier to suppress a parasitic oscillation that often shows up in such circuitry. The resistor has no effect upon the operation of the amplifier stage. Figure 15 OBLIQUE VIEW OF TRANSCEIVER CHASSIS The 2(26 power amplifier tube is mounted in a horizontal position, supported by bracket at rear of the chassis. Chassis is perforated below tube to permit passage of air around tube. Pi- network components are in front of tube cap. Antenna coaxial receptacle is at rear of chassis on bracket. Construction of multiple front panel may be seen in this view. Meter Transceiver 541 A triode -pentode (6CG8) is used as the first mixer stage. The triode section operates as a "hot plate" oscillator 2050 kc. below the signal frequency. Grid injection is used to the pentode mixer section. Parallel padding capacitors are switched across the mixer and oscillator circuits in order to tune the 15 meter band. A 6BE6 pentagrid tube is employed as a second mixer from 2050 kc. to 265 kc. The #1 grid acts as the anode of a cathode feedback crystal oscillator, with the degree of feedback controlled by a capacity bridge placed between #1 grid, cathode and ground. A single 6BÁ6 provides sufficient i.f. gain at 265 k.c., and two transformers produce excellent "skirt" selectivity and adjacent signal separation. The r.f. stage, the second mixer, and the i.f. stage are all controlled by the a.v.c. circuit, operating from one -half of a 6AL5 tube. The second diode section of the 6AL5 acts as the second detector and automatic noise limiter. The a.n.l. circuit has a very low distortion level, and is in the circuit at all times. A 6ME -10 miniature "magic eye" tube serves as a signal strength indicator, operating from the a.v.c. line of the receiver www.americanradiohistory.com THE RADIO Receivers and Transceivers 540 V --O A-P AMP. MULTIPLIER VIAL 05C. ()MC) (21ORjCMC) MIC VOL. B.6300 ei 1'O V. V. Figure 14 BLOCK DIAGRAM OF TRANSCEIVER External power supply is used, so transceiver may be operated from a.c. supply or from mobile power pack. Tuning adjustments are accomplished by means of 6E5 miniature "magic eye." The "eye" tube shown is the imported type 6ME -10 (Concord Electronics Co., 809 No. Cahuenga Blvd., Los Angeles 38, Calif.). The FM -type EM -84 may be substituted. has arisen for a compact transceiver that will work well either in the car or at the home station. The unit described in this section has been designed to meet this need. This compact transceiver package covers the 10 and 15 meter bands, and employs a stable superheterodyne receiver and a 20 watt a.m. transmitter. The transmitter may be either crystal controlled, or driven by the internal v.f.o. A 10 watt audio system operates from a crystal microphone and provides 100% modulation of the transmitter. During reception the audio stages deliver sufficient power to drive an external speaker well above the noise level of the automobile. Small enough to fit comfortably under the dash of today's car, the transceiver delivers a well modulated signal at a maximum plate power load of 300 volts at 170 milliamperes and 150 volts at 20 milliamperes. Transceiver Circuit A block diagram of the trans ceiver circuit is shown in figure 14. Twelve tubes are employed, three in the transmitter section, three in the audio portion, and six in the receiver section. Change-over from receive to transmit is accomplished by a four -pole, two position switch (S1), mounted in the upper left corner of the front panel (figure 13). One section of this switch (Six) removes the plate and screen voltage from the transmitter amplifier stage, another section (S).p) transfers the low voltage from the transmitter exciter stages to the receiver circuits, a third switch section (Sl.D) activates either the speech amplifier or the loud speaker jack, and the fourth section (S)_B) sensitizes the "magic eye" indicator tube for reception. The receiver portion employs a double conversion circuit to achieve maximum image rejection with adequate selectivity. A 6DC6 www.americanradiohistory.com HANDBOOK 10 -15 fairly broad as is the r.f. stage tuning although the antenna trimmer will have to be repeaked when going from one end of the 80 meter band to the other. However, if the mixer trimmer capacity has to be changed at the low end of a band to obtain an increase in S -meter reading, it means that the coil will have to be altered. If the trimmer has to be increased in capacity, it indicates that more inductance is needed and the tap will have to be moved farther up on the coil. Conversely, if the trimmer has to be decreased in capacity, less inductance is needed and the tap is moved down on the coil. The Erie trimmer capacitors pass from maximum to minimum capacity in 180 degrees of rotation and are at minimum capacity when the lettering on the cap is adjacent to the mounting bolts. The final dial calibration must be made with a bottom shield on the chassis temporary piece of screen wire is satisfactory in order that the calibration will be correct when the receiver is placed in the cabinet. 100 kc. points are marked off on the dial from harmonics of the crystal calibrator and in between points may be marked off with dividers since the dial is fairly linear. The dial scale is removed for inking the calibration points and when permanently reinstalled, it is covered with a 1/16 inch piece of clear polystyrene sheet the same size as the scale. This will keep the dial clean and prevent warping of the paper scale. A permanent pointer is made from a long scrap of polystyrene sheet with an inked -a - Meter Transceiver 539 line scribed down the center of the pointer. It can be shaped to fit over the planetary drive by holding the plastic under hot water until it is soft enough to be bent to shape. The front panel is fastened to the chassis by means of the hexagonal nuts holding the bandswitch and toggle switches. Another 1/16 inch sheet of polystyrene or plexiglass is cut to fit over the dial opening with the cutout for the shaft of the planetary drive allowing clearance for the tuning knob. Lettering decals are used to mark the controls. Any well filtered power supply that delivers about 250 volts at 80 ma. is suitable for the receiver. For 6 volt operation terminal 4 on the power plug is jumpered to terminal 2 (ground) and power is applied to terminal 3. For 12 volt operation terminal 3 is left open and 12 volts applied to terminal 4. 27 -4 A Compact Transceiver for 10 and 15 Meters Regardless of "conditions" and the sunspot cycle, the 10 and 15 meter bands are exceedingly popular with a large group of amateurs. Many stations on these bands employ low power, and the amateur using a low power transmitter, inexpensive receiver and modest antenna suffers no great handicap. In addition, the growth of mobile operation on these bands has been rapid and the need .................. ... .......................41 . ... .... 09400001,160000.0000000.0001000 00041"0004108000041000000061,100410000 ................................. ..0..00....41..0.......0.000...... 4000060.0.41.0410.0000.000000.0000000 0000.041414100041414141000000000000 ....41.41...........41........0.0..... ........................... 41.41.000041414141410.0000041..414141000000.0. 0041.0.41414141.4100000.....0.0.....0 0.4141414141....0...00...41410.00000.00. 41041414141414100000.141000000410000004104100 0...410.0414141.0.41.041.0.0.00000...41 ...041.41........ 4141.......41.4141.4141.....ci .. .. ...........41.......... .. .......4141............. 04104.40004100000.0000000000000000.0 Figure 13 POCKET -SIZE TRANSCEIVER! This miniature transceiver is designed for top-performance on the 21 and 28 Mc. amateur bands. Receiver section utilizes double conversion for suppression of images and for maximum selectivity. Modulated amplifier stage of transmitter employs a 2E26. Power supply and speaker are contained in auxiliary cabinet shown sitting atop transceiver. www.americanradiohistory.com 538 THE RADIO Receivers and Transceivers The next step in the assembly of the receiver is to mount the coil partitions with the attached switch sections. The shaft is connected to the switch index (previously mounted on the front of the chassis) with a 1/4 -inch shaft coupling and the switch index is rotated so that the switch contacts are in proper alignment. The nut holding the index can then be tightened down to hold it in place. The r.f. coils are wound as shown in the coil table (figure 11) and the two grid coils (L1 and L3) can be wired in without further attention since they tune separately with the antenna trimmer. The third set of contacts on the grid coil switch wafer (SW1.c) are used to switch in the additional 680 ohm cathode resistor used to limit the excessive gain developed on the 80 meter band. This prevents overloading the triode mixer with a strong signal. The mixer and oscillator coils are next wound and installed and the r.f. section is ready to be tuned up. Coil Assembly proceeding with the alignment of the oscillator and mixer coils, a dial scale is made up from stiff paper or white cardboard and fastened to the dial back plate. The semicircular scales are made with a compass using black india ink. A temporary pointer is made of light aluminum or plastic scrap and is attached to the planetary drive dial plate by means of the small screws holding the dial plate. The section of the pointer extending over the scale is cut in half along the center line so that only half of the pointer remains to be used as a guide line for marking off the calibration points on the dial. A rough alignment of the oscillator and mixer coils can be made with a grid dip meter using the tuning range data given in the coil table. The frequency of the oscillator circuit is higher than the frequency of the mixer circuit by 3000 kc. (the frequency of the i.f.) except in the case of the 15 meter band where the oscillator frequency is lower than the mixer frequency by 3000 kc. As a starting point, the tuning capacitor is set at almost full mesh (near maximum capacity) and each oscillator coil is set at its lowest frequency by adjusting its associated trimmer to maximum capacity. The mixer coils can be set in the same fashion on Receiver Calibration Before the low frequency edge of each amateur band. The low band edges can be found by having the receiver turned on for operation and applying a signal to the antenna input from a signal generator or your transmitter v.f.o. The antenna trimmer will have to be peaked for each band. Once the low band edges have been found, the rest of the alignment and dial calibration is easily done using the built in 100 kc. calibrator. A short piece of wire is clipped to the plate of the crystal calibrator tube and brought near the antenna coils to get a fairly strong signal to work with. A crystal harmonic should fall at the low end of each band at the point on the dial found previously with the external signal. With this as a starting point, the tuning capacitor can be rotated over the dial range and 100 kc. points counted off to check the coverage of each band on the dial. The coil and capacitor combination given in the coil table is designed to spread each band over almost the entire 180 degrees of the dial using a 15 µµfd. tuning capacitor. If an entire band cannot be covered, the turns on the particular oscillator coil must be squeezed together slightly to increase the inductance and the alignment procedure repeated by resetting the oscillator trimmer so that the dial pointer will align with the original calibration point at the low end of the band. If the bandspread is insufficient, the coil turns must be spread to decrease the inductance. In the case of the 80 meter band it may mean removing one or two turns from the oscillator coil. When the band coverage has been set by the above adjustment of the oscillator coils and associated trimmers, the mixer coils are adjusted for proper tracking. Using the same harmonics from the 100 kc. calibrator set the tuning dial at the high frequency end of the band, peak the antenna trimmer for a maxiTracking Adjustments mum reading on the S-meter and likewise peak the trimmer capacitor on the mixer coil. Rotate the tuning capacitor to the low frequency end of the band and again adjust the same mixer trimmer for a maximum reading on the Smeter. If no improvement can be made in the meter reading, the mixer is tracking with the oscillator and no further adjustment need be made. The 80 and 40 meter mixer tuning is www.americanradiohistory.com HANDBOOK Bandpass- Filter Receiver 537 Figure 12 iSWiTC 5.6n SS Mill PLACEMENT OF MAJOR COMPONENTS BENEATH THE NOES' 120 UUF, O o 0 O0A2 I 5W3 6Ue 0 11 Le 6BZ6 o Ol C+ O 1. I e 6005 'lo 01: SW eue 0 BFO PITCH 0 _OJ The 6BZ6 r.f. tube socket Tz eTe o i° LO_ located so that the rear wall of the coil compartment passes over the center of the socket, isolating the input and output circuits. The 6U8 mixer socket is mounted in the some fashion with the mixer pins (I, 8, and 9) falling within the coil compartment and the oscillator pins (2, 6, and 7) placed in the oscillator coil area of the chassis. Bandchange switch segments are mounted to the walls of the coil compartment and are driven by the switch index affixed to the front wall of the chassis. The antenna trimming capacitor and b.f.o. pitch capacitor are mounted to extension "ears" on the rear wall of the coil compartment. is I :10 60.16 LO CHASSIS o 808 `iYi 5w2 L9 \--SHAFT COUPLING _0J of the ceramic trimmer capacitors and the coil leads are connected to the same lugs, with a heavy wire going to the respective terminals on the band switch. There is a continuously shorting deck on the oscillator switch section that picks up the unused coils and shorts them all together to prevent any absorption loss from occurring in the higher frequency coils. The two 3000 kc. i.f. transformers are made from standard 1500 kc. units by removing 5 feet of wire from each winding of the transformer. The transformers shown are J. W. Miller # 13 -W 1, but other makes of transformers could also be altered to tune to 3000 kc. The proper frequency is easy to check with a grid dip meter. The b.f.o. coil is made the easy way by using a broadcast -type "vari- loopstick." Twelve inches of wire is removed from the coil and an additional 18 inches is unwound to make the cathode tap and is then rewound back on the coil. The end of the rewound section of the coil will be the ground end. A padding capacitor of 68 µµfd. is placed across this coil and the modified " loopstick" is mounted in a shield can to match the i.f. transformers. The slug of the coil tunes the circuit to the exact frequency and the variable pitch control capacitor between cathode and ground provides about 3 kc. variation each side of zero beat. Preliminary Checking and Adjustment Most of the wiring can be done before the coil partitions and the r.f. coils are installed. At this point the tubes can be put in the sockets (figure 12) and power applied for a preliminary check on the i.f. and audio circuits. Tuning the i.f. channel is a simple matter because the center frequency is determined by the bandpass filter. A low level 3000 kc. signal from a signal generator or grid dip meter is applied through a capacitor to the input of the filter and the slugs of the i.f. transformers are adjusted for maximum signal reading on the S- meter. The S -meter is adjusted with the "zero" control to read zero with no signal and the meter sensitivity is determined by the value of the meter ground resistor. The value of 47K used with the 0 -1 ma. meter will give full scale deflection with a very strong local signal. www.americanradiohistory.com THE RADIO Receivers and Transceivers 536 near that part of the circuit in which they are used to allow short leads and to facilitate wiring (figure 10) Terminal boards can be made up from one inch wide strips of fiberglass sheet or phenolic material using soldering lugs and rivets for the terminals. A plain piece of the same material is placed between the terminal board and the side of the chassis and the two pieces are fastened to the chassis with 4 -40 hardware. The trimmer capacitors for the mixer coils are mounted on the coil partition so that their terminals extend into the mixer coil section adjacent to the proper coil. 4 -40 tapped holes in the lip of the partition allow the trimmers to be fastened directly to the partition. These trimmers can be wired to the band switch before the partition is mounted to the chassis, and the leads of the coils may be soldered to the trimmer capacitor terminals when they are installed. Notice that one terminal of the trimmer capacitors returns to a ground strap allowing them to be . tuned with a metal screw driver without shorting the B -plus voltage appearing on the mixer coils. The bottom leads of the mixer coils return to a common bus wire run near the chassis and terminating on an insulated tie point near the 80 meter coil. Shielded wire (phonograph pickup type) is used in the audio circuit to bring the leads from the 6T8 socket to the volume control, but all r.f. wiring is unshielded, short and direct. B -plus leads, S -meter wiring and the a.v.c. and r.f. gain control wires pass around the inside edge of the chassis where they are out of the way. The disc -type bypass capacitors are mounted right at the tube sockets with short leads. The metal center posts of the r.f. and i.f. tube sockets are grounded and the cathode bypass capacitors are positioned to cross the center of the sockets to further isolate the input and output circuits of the tubes. The silver mica padding capacitors in the oscillator circuit mount directly on the lugs Figure 11 COIL DATA All r.f. coils are wound on 138" lengths of !'2" diameter polystyrene rod in a space of 2/4" except coil Li which is close wound in 9 16" space. The antenna coils are wound at the ground end of the grid coils and spaced 1 16" below the grid coil. All wire is plain enamel in the sizes shown. Holes are drilled through the forms to fasten the end of the coils and the forms are tapped for 4 40 bolts at the bottom ends to attach them to the chassis. The tcp point is the number of turns from the bottom (B -plus) end of the coil. BAND COIL TUNING RANGE 80 L1 Grid 3500 -7300 Ant. Mixer 3500 -4000 Osc. 6500 -7000 45 #30 17 30 30 26 Mixer 7000 -7300 Osc. 10 -10.3 Mc. 35 L11 L,; Grid 14 -30 Mc. L L Ant. Mixer 13 10 L., L1_19 40 20 15 10 L, TURNS 20 14 Lti Mixer Osc. 18 -18.5 Mc. 7 1/2 L, Mixer 28 -30 Mc. 9 L1d Osc. 31 -33 Mc. B.f.o. coil BC 51/2 - 6 7 L1s -21.5 Mc. 6 16 14 -14.4 Mc. Osc. 17 -17.4 Mc. L1 WIRE 80 37 L12 21 TAP "vari -loop tick" 6 7 (see text) www.americanradiohistory.com TRIMMER FIXED PAD µµtd. ppfd. C-1 8 -50 4 -30 26 20 8 -50 20 30 20 20 C-1 20 20 8 -50 20 20 8 -50 4 -30 8-50 4-30 4 -30 4 -30 None 85 120 110 30 HANDBOOK Bandpass -Filter Receiver SW6 0 535 POWER PLUG Figure 10 o vTCRMI TERMINAL BOARD UNDER -CHASSIS REAR APRON LAYOUT AND PLACEMENT OF TERMINAL BOARDS Terminal boards are NAL. BOARD TERMINAL BOARD I I COIL COMPARTMENT I L I J LEFT SIDE RIGHT SIDE the dial which is 53/4" long and /8" high. It has 3/8" lips and takes the form of a shallow pan. Two holes are drilled in the bottom lip of the pan to fasten it to the chassis flush with the front edge. The center of the pan is in line with the center of the chassis and the capacitor shaft. Tapped 6 -32 holes in the chassis make it easy to mount the dial pan. The center hole for the planetary drive is found by sliding the back plate of the dial up against the capacitor shaft and marking the location of the hole. The drive mechanism is then positioned on the back plate and bolted to it. After these parts are permanently mounted, a couple of braces are affixed from the back plate of the dial to the chassis so that the whole dial assembly and tuning capacitor are held rigid (figure 8). The Coil Assembly Light sheet aluminum is used to make the under -chassis partitions that separate the coils and act as mounting plates for the bandswitch sections. The rear partition holding the antenna switch section, the antenna trimmer and the b.f.o. capacitor measures 71/2" x 17/8" with 1/4 -inch lips bent at right angles to the top and bottom (figure 9) . The lips provide stiffening and permit easy mounting to the chassis. The front partition holding the oscillator and mixer switch sections measures 51/2" x 1 %s ", with the same 1/4 -inch lips top and bottom. The mixer coil trimmers are mounted on the lip and do not project beyond the depth mounted on the left, right, and rear side walls of the chassis. A piece of phenolic material placed beneath the board insulates the terminals from the chassis. R.f. bypass capacitors are mounted directly to the pins of the tube sockets. of the chassis. Two side pieces attach to the partitions to hold them rigid. Threaded holes are cut in the bottom lips of the partitions so that they can be fastened to the chassis with 4 -40 self- tapping screws. Cutouts are made in each partition to clear the r.f. and oscillator tube sockets and two 1/4-inch feed -through holes are drilled in one side partition for the B -plus and plate leads to the r.f. stage. The bandswitch wafers, SW1, SW2, and SW3 are mounted with bolts and spacers on the coil partitions on a center line directly below the tuning capacitor. The antenna trimmer capacitor and the b.f.o. pitch capacitor are mounted on the rear partition three inches to each side of the band switch center line. Fiber extension shafts bring the controls up to the front panel. A flat sided fiber shaft is passed through the band switch sections and is connected to the switch index with a metal shaft coupling. The switch index is mounted on the front apron of the chassis. The location of the holes to be drilled in the panel is found by placing the drilled chassis against the panel and marking the holes from the inside of the chassis. The large cutout for the dial and the S-meter can be made with a fine toothed coping saw. Wiring the Three small phenolic terminal boards are used to mount most of the miscellaneous resistors and the audio coupling capacitors. The boards are bolted to the side of the chassis Receiver www.americanradiohistory.com 534 THE RADIO Receivers and Transceivers chassis by 6 -32 bolts in the space between each capacitor. The ground terminals of the coils, trimmers, and padding capacitors attach to ground lugs affixed to the chassis at the ground bolt of the trimmers. The tuning capacitor is mounted on a small L- shaped bracket and is inverted to place the terminals and ground strap closer to the chassis. The ground strap on the capacitor projects through a hole in the chassis and makes connection to the ground terminal underneath the chassis. The tuning capacitor is positioned along the center line of the chassis at a distance from the front panel determined by the length of the dial drive mechanism. For smooth tuning the planetary drive must be lined up carefully with the shaft of the tuning capacitor. The drive mechanism mounts on the back plate of The Dial Figure 9 UNDER -CHASSIS VIEW OF RECEIVER The general layout of components beneath the chassis is shown in this view. The bandswitch passes through the central coil comportment. Each switch segment is mounted to a wall of the compartment. The two antenna coil forms are mounted behind the coil compartment with the primary windings connected to the antenna receptacle by a short length of coaxial line. To the right is the r.f. stage tuning capacitor, and to the left is the b.f.o. pitch capacitor. The bank of mixer coils is centered in the compartment and the various trimming capacitors are mounted on the top front flange of the compartment to facilitate adjustment. Note that the rear wall of the compartment passes across the socket of the r.f. stage, thus isolating the input and output circuits of the tube. The bank of oscillator coils is mounted between the compertment and the front wall of the chassis, with the oscillator switch mounted to the outside wall of the compartment. All wiring is short and direct, and most small bypass capacitors are mounted directly to the tube socket pins. Resistors are mounted on terminal boards placed on the walls of the chassis. ! www.americanradiohistory.com Bandpass -Filter Receiver up for in the following i.f. stage using standard transformers. The bandpass filter requires no special coupling circuits and is symmetrical as viewed from its terminals in other words, either terminal may serve as input or output at a nominal impedance of 4700 ohms. Mixer plate voltage may be applied directly to the filter since it is tested at a voltage far in excess of any value normally encountered in receivers. The gain control is in the cathode of the i.f. stage as well as the r.f. stage for more effective control of the over -all gain of the receiver. The second 6BJ6 i.f. tube has its screen voltage regulated for more effective operation of the bridge -type S -meter in its plate circuit. - The Detector and The detector and audio stages are conventional and the noise limiter is a series diode (part of the 6T8 tube) with a fixed threshold level that does a good job of limiting noise without causing apparent distortion on phone signals. A switch is included to disable the limiter when it is not needed. The audio volume control is isolated from d.c. potentials by coupling capacitors to eliminate the tendency of these controls to become noisy Audio Stages when used in current carrying circuits. A standby switch opens the cathode of the 6AQ5 stage silencing the receiver, but other methods can be used such as cutting the B -minus of the separate power supply or relay switching the B -plus of a mobile power supply. Receiver The 81/2" x 11" x 2" chassis size is just right for building this receiver and there is plenty of room for all the components without crowding, even if the exact parts specified are not used. Location of the major components is shown in the photographs and the layout follows the circuit diagram (figure 7) , passing around the chassis with the r.f. section taking up most of the center area. It is a good idea to make paper templates for the band pass filter and the i.f. transformers, marking the drilling holes on the chassis from the templates. The various tube sockets should be oriented so that grid and plate leads do not have to cross over the sockets. Oscillator trimmer capacitors are mounted on top of the chassis in a line above the oscillator coils with the capacitor leads projecting through 1/4 -inch holes to the under side of the chassis. The oscillator coils can then be mounted under the Construction Figure 8 REAR VIEW OF BAN DPASS- FILTER RECEIVER The audio stages, 100 kc. calibrator crystal, and b.f.o. are located along the left edge of the chassis. Intermediate am- plifier stages 533 pass across the back of the chassis, with the 3 Mc. bandpass crystal filter and voltage regulator tube at right. Oscillator alignment capacitors are placed directly behind the homemade dial, with the main tuning capacitor centered on the chassis. The r.f. and mixer tubes are at the center of the chassis. Along the rear apron of the chassis are (I. to r.): audio jacks and a.n.l. switch, antenna receptacle, and power plug. The 5 -meter "zero"-set potentiometer is placed atop the chassis, between the 100 kc. crystal and the first i.f. tube. 411 www.americanradiohistory.com II N Ia N J VI J a ; ] - U O y0 3aó VI .."-i'' 40w-1I' . ; uT 0--10 U 11 x ú n- ---H z OQQfJ y- o Q(44f z u www.americanradiohistory.com E$ HANDBOOK Bandpass -Filter Receiver plug and antenna connector. The most -used controls are on the front panel. The tuning dial is a very smooth planetary type having a long plastic pointer which extends over a calibrated scale. Indirect lighting of the dial is provided by two panel bulbs recessed at the edge of the dial scale. A direct reading S -meter is connected in the plate circuit of the last i.f. amplifier tube, and a 100 kc. calibrator is included. Receiver Circuit The tube lineup consists of a high gain 6BZ6 semi remote cutoff r.f. amplifier; a 6U8 triode mixer and oscillator; two 6BJ6 i.f. amplifiers; a 6T8 detector, noise limiter and audio amplifier; and a 6AQ5 audio output stage. A second 6U8 is used for a combined beat oscillator and crystal calibrator, and an 0A2 serves as a voltage regulator. The circuit is conventional with several simplifications that make for less work and easier construction, to say nothing of fewer parts (figure 7) . The main d.c. supply voltage to the plates and screens of the tubes is divided by resistors to eliminate coupling through a common voltage source so extra decoupling networks are not required. A voltage regulator tube stabilizes the oscillator and screen voltages as well as the voltage on the b.f.o. Bandswitching for the five amateur bands is accomplished with only three switch wafers (SWIA,B,c, SW2A_B, and SW3) and the main tuning dial is used only for the oscillator circuit (C2B) and plate coil of the r.f. stage (C.,A) . This calls for only a small inexpensive two gang capacitor. Only two coils (L1 and L3) are used in the r.f. grid circuit to cover five bands. Coil L1 covers the 80 and 40 meter bands and coil L3 covers the 20, 15 and 10 meter bands, trimmed by the 100 µµEd. antenna capacitor, C1. The r.f. tuning is fairly broad and the trimmer only requires resetting when going from one end of a band to the other. The plate circuit of the r.f. amplifier (C2A, L5.0) is resonated using separate coils for each band which are preadjusted to track with the oscillator tuning. Except for the 80 meter band, the plate coils are tapped for proper oscillator tracking, eliminating the need for extra series or padding capacitors required with other tracking methods. The oscillator 531 circuit (C2B,L10 -L14) uses single winding coils in a Colpitts arrangement utilizing large padding capacitors which serve the dual purpose of stabilizing the oscillator as well as providing proper bandspread for the tuning ranges. The oscillator tuning capacitor C2B is as small in capacity as can be used to cover the desired tuning range. The oscillator circuit is designed as if it were going to be used for a v.f.o. in a transmitter which calls for mechanical rigidity and use of short leads, a ceramic switch and tube socket, silver mica capacitors, and solidly mounted coils. The coils are wound on polystyrene rods and are bolted to the chassis. Directly above each coil (on top of the chassis) are the ceramic trimmer capacitors (C4 -Cg) used to adjust the tuning range (figure 8) . The capacitor lugs project through holes drilled in the chassis and fall adjacent to their respective coils and switch contacts. The trimmer capacitors have a negative temperature coefficient, and the combination of fixed silver mica padding capacitors and the negative compensating characteristics of the adjustable trimmers tends to stabilize the oscillator frequency with respect to temperature changes. Ceramic capacitors, together with the small, rigid plates of the tuning capacitor make the oscillator almost immune to vibration. - The Mixer Stage A triode mixer is not commonly used in band switching receivers but its low internal noise and low plate resistance make it ideal for working into the low impedance of the crystal bandpass filter. The injection voltage from the oscillator is fed directly into the cathode of the triode section of the 6U8. Although the injection voltage varies from one band to the next, the value is not critical and is sufficient on all bands. A v.t.v.m. reading at the grid of the oscillator (pin 2) will show between and volts for proper operation. A 5600 ohm resistor is placed across the 80 meter oscillator coil (L10) to reduce the injection voltage on this band to the proper level. -5 -8 The I.F. Amplifier The crystal bandpass filter (Blackhawk Engineering Co.) has a maximum insertion loss of only 3 db. This loss is more than made www.americanradiohistory.com 530 in the collector circuit. Optimum collector load for the 2N217 is approximately 500 ohms, and the 2N217 develops a maximum audio signal of 75 milliwatts at this load impedance. Transformer T, matches the transistor circuit to the 12 ohm miniature loudspeaker. The receiver draws a maximum signal current of 11 milliamperes from the 9 -volt battery supply. If no external antenna is used, the receiver should be moved about to orient the "loopstick" coil L, for best pickup of each individual broadcast station. Adjacent channel interference can often be eliminated by careful rotation of the set to "null out" the offending signa:. Ample loudspeaker volume will be obtained from local stations without the use of an external antenna. 27 -3 THE RADIO Receivers and Transceivers An Inexpensive Bandpass- Filter Receiver A very selective high performance amateur band receiver can be built by using a high frequency crystal bandpass- filter in the i.f. system. Selectivity and image rejection are accomplished without the complex circuitry and elaborate construction required in a dual conversion receiver to obtain the same results. The intermediate frequency used in this receiver is 300 kc., yet the bandwidth is only 3 kc. at 6 db down and 12 kc. at 60 db down with sharp skirt selectivity. The receiver covers all amateur bands between 10 and 80 meters. To supplement this efficient i.f. system, the entire receiver has been designed towards the goal of simplicity both in circuitry and mechanical construction without sacrificing anything in performance or leaving out any controls needed in a communication receiver. The receiver is built on a standard size chassis with a compact perforated cabinet suitable for use in the shack, or in the car as a first rate mobile receiver (figure 6) . The power supplies are built as a separate unit for this purpose and the tube filaments are wired in a series parallel configuration so that either a 6 or 12 volt d.c. supply may be used. The speaker is external and the seldom -used noise limiter switch and headphone jack are on the rear apron of the chassis, along with the power Figure 6 HIGH PERFORMANCE AMATEUR BAND RECEIVER MAKES USE OF 3 MC. CRYSTAL I.F. FILTER This simple, easy to build receiver achieves a near ultimate in selectivity and sensitivity without the complex circuitry of a dual conversion receiver. The bandspread dial is made up from an inexpensive vernier unit, to which a celluloid pointer has been attached. The dial opening is covered with a thin sheet of lucite, held in position with 4 -40 sheet metal screws. Directly below the tuning dial is the band switch with Vie antenna trimmer, the a.v.c. and b.f.o. switch, and the i.f. gain control to the left. Above these is the S- meter, mounted to the rear of the panel. To the right are the b.f.o. pitch control, the standby and calibration switches, and the audio gain control. The receiver sits on four rubber feet to prevent the operating table from being marred by the metal case. The front panel is attached to the receiver chassis, and the ventilated cabinet is bolted to the rear of the chassis. Receiver size is only 11" x 6 ". www.americanradiohistory.com HANDBOOK 529 Circuitry and 27 -1 Components It is the practice of the editors of this Handbook to place as much usable information in the schematic illustration as possible. In order to simplify the drawing the component nomenclature of figure 1 is used in all the following construction chapters. The electrical value of many small circuit components such as resistors and capacitors is often indicated by a series of colored bands or spots placed on the body of the component. Several color codes have been used in the past, and are being used in modified form in the present to indicate component values. The most important of these color codes are illustrated in figure 2. Other radio components such as power transformers, i -f transformers, chokes, etc. have their leads color -coded for easy identification as tabulated in figure 3. 27 -2 A Simple Transistorized Portable B -C Receiver Illustrated in figures 4 and 5 is an easy to construct two transistor portable broadcast receiver that is an excellent circuit for the beginner to build. The receiver covers the range of 500 kc. to 1500 kc. and needs no external antenna when used close to a high power broadcasting station. An external anten- na may be added for more distant reception. The receiver is powered from a single 9 -volt miniature transistor battery and delivers good speaker volume, yet draws a minimum of current permitting good battery life. Circuit Operation of the receiver may be understood by referring to the schematic diagram of figure 5. The tuned circuit L1 -C, resonates at the frequency of the broadcasting station. A portion of the r -f energy is applied to the base of the 2N112 p -n -p type transistor. A tapped winding is placed on coil LI to achieve an impedance match to the low base impedance of the transistor. Emitter bias is used on this stage, and the amplified signal is capacitycoupled from the collector circuit to a 1N34 diode rectifier. The rectifier audio signal is recovered across the 2K diode load resistor, which takes the form of the audio volume control of the receiver. The diode operates in an untuned circuit, the selectivity of the receiver being determined by the tuned circuit in the r -f amplifier stage. The audio signal taken from the arm of the volume control (R1) is applied to the base of the 2N112 r-f amplifier which functions simultaneously as an audio amplifier stage. The amplified audio signal is recovered across the 2K collector load resistor of the 2N112, and is capacitively coupled to the base of a 2N217 p -n -p audio transistor. This stage is base and emitter biased, having the output transformer Description Figure 5 INTERIOR VIEW OF TRANSISTOR RECEIVER The speaker and output transformer are mounted at the left of the Masonite chassis. Top, center is the "loopstick" r-f coil, and directly to the right is the 10 millihenry r -f choke in the collector lead of the 2N112 transistor. Battery Is at lower right. www.americanradiohistory.com 528 THE RADIO Receivers and Transceivers Transformer, choke and coil windings may be damaged by incorrect wiring of the high -volt- FIGURE 3 COMPONENT COLOR CODING age leads. POWER TRANSFORMERS PRIMARY LEADS The problem of meeting and overcoming such obstacles is just part of the game. A true radio amateur (as opposed to an amateur broadcaster) should have adequate knowledge of the art of communication. He should know quite a bit about his equipment (even if purchased) and, if circumstances permit, he should build a portion of his own equipment. Those amateurs that do such construction work are convinced that half of the enjoyment of the hobby may be obtained from the satisfaction of building and operating their own receiving and transmitting equipment. BLACK IF rAPPEO COMMON BLACK TAP BLACK /YELLOW END BLACK HIGH VOLTAGE WINDING / RED RED CENTER -TAP RED /YELLOW RECTIFIER FILAMENT WINDING- YELLOW YELLOW /BLUE CENTER -TAP FILAMENT WINDING N GREEN 1 GREEN /YELLOW CENTER -TAP FILAMENT WINDING N BROWN 2 CENTER -TAP FILAMENT WINDING BROWN N /YELLOW SLATE 3 SLATE /YELLOW CENTER -TAP The Transceiver I-F TRANSFORMERS PLATE LEAD BLUE 0+ LEAD RED GRID (OR DIODE LEAD ) A -V -C (OR GROUND) GREEN LEAD BLACK AUDIO TRANSFORMERS PLATE LEAD (PR/.) BLUE OR BROWN (PR /. ) GRID LEAD (SEC.) GRID RETURN (SEC.) RED BY LEAD GREEN OR YELLOW BLACK circuits. If possible, the wiring should be checked by a second party as a safety measure. Some tubes can be permanently damaged by having the wrong voltages applied to their electrodes. Electrolytic capacitors can be ruined by hooking them up with the wrong voltage polarity across the capacitor terminals. A popular item of equipment on "five meters" during the late "thirties," the transceiver is making a comeback today complete with modern tubes and circuitry. In brief, the transceiver is a packaged radio station combining the elements of the receiver and transmitter into a single unit having a common power supply and audio system. The present trend toward compact equipment and the continued growth of single sideband techniques combine naturally with the space-saving economies of the transceiver. Various transceiver circuits for the higher frequency amateur bands are shown in this chapter. The experimenter can start from these simple circuits and using modern miniature tubes and components can design and build his complete station in a cabinet no larger than a pre -war receiver. EXTERNAL ANTENNA 10H .002 B20 0.0.5 C o.5 100 LFD + is Rr B HIVOLUNE A30 2 n o.S 100LF0 + 15 BUS Figure 4 SCHEMATIC OF TRANSISTOR BROADCAST RECEIVER -volt transistor battery. RCA VS -300 -9 -365 Nµfd. Lafayette Radio Co. MS -214, or Allied Radio Co. 61H -009 "Loopstick" coil. Lafayette Radio Co. MS -166 loudspeaker, 12 -ohm voice coil. Lafayette Radio Co. SK-39 T,-500 ohm pri., 12 ohm sec. Transistor transformer. Thordarson TR -18 C, L,- Transistor LS -3" www.americanradiohistory.com 527 times such a comparison is surprising. When the builder has finished the wiring of a receiver it is suggested that he check his wiring and connections carefully for possible errors before any voltages are applied to the experimenter's instinct, even in those individuals owning expensive commercial receivers. These lucky persons have the advantage of comparing their home -built product against the best the commercial market has to offer. Some- STANDARD COLOR CODE- RESISTORS AND CAPACITORS AXIAL LEAD RESISTOR BLACK _ M' l -.'G1 -- TOLERANCE MULTIPLIER TOLERANCE i 00 MULTIPLIER 3 YELLOW 4 TOLERANCE GREEN S 5 .000 0.000 00.000 000.000 0,000.000 00,000.000 000.000,000 IS e VIOLET 7 7 GRAY e e 9 WHITE 9 COEFF CURE COEFFICIENT IJIISIII ('' rT, COEFF L'JI)aL:L4aIl1 I- TOLERANCE TOLERANCE MULTIPLIER LTC MULTIPLIER MULTIPLIER BY -PASS COUPLING CERAMIC CAPACITOR I 1.11 OIIM1111I- IGURE LIST FIGURE AXIAL LEAD CERAMIC CAPACITOR CAPACITY TEMP. COEFF. CAPACITY -MULTIPLIER I TEMPERATURE 5- ROTRADIAL LEAD CERAMICCAPACITOR EXTENDED RANGE TC CERAMIC HICAP RADIAL LEAD (BAND) RESISTOR Q1111 I 2 1ST FIGURE TOLERANCE CAPACITY 0 4 ¡ MULTIPLIER II 3-DOT 5 -DOT NONE 2 3 RADIAL LEAD DOT RESISTOR %1I DOT COLOR MULTIPLIER 1 BLUE WIRE-WOUND RESISTORS NAVE 1ST DIGIT BAND DOUBLE WIDTH. DISC CERAMIC RMA CODE THIRD RING 0 0 BROWN RED ORANGE 2ND SIGNIFICANT FIGS I STa SECOND RING END COLOR SECOND FIGURE INSULATED FIRSTRING BODY COLOR UNINSULATED FIRST FIGURE COLOR DROWN- INSULATED BLACK - NON -INSULATED V(OPT `E O I NI! TOLERANCE MULTIPLIERS TOLERANCE MULTIPLIER MOLDED MICA TYPE CAPACITORS RMA 3 -DOT (OBSOLETE) CURRENT STANDARD CODE 1 ST L SIGNIFICANT FIGURE RATED 300 V.D.C. ± BLACK(JAN) ® CLASS 1srL ( ¡ 2ND ' JAN o eljAULTIPLIER MULTIPLIER CODE i____%7 SIGNIFICANT FIG. "-TOLERANCE - RMA 6-DOT (OBSOLETE) RMA 5 -DOT CODE (OBSOLETE) sIa FIGURE WORK. TOLERANCE VOLT MULTIPLIER FRONT In t /9OB RMA EMULTIPLIER WORK. VOLT. REAR WORK. TOLERANCE f ND SIG. FIG. 1ST I , CLASS TOLERANCE¡F A 4s,L 2ND WHITE (RMA)",....te BUTTON SILVER MICA CAPACITOR 20 % TOL. MULTIPLIER 1ST ,qp l MULTIPLIER 3RD DIGIT RMA 4 -DOT (OBSOLETE) } SIG. FIGURE WORK. VOLTAGE J MULTIPLIER MULTIPLIER L L TOLERANCE TOLERANCE 1ST DIGIT 2ND DIGIT 2NDJ FIGURE 157 j SIG. WORKING VOLTAGE BLANK VOLT. MOLDED PAPER TYPE CAPACITORS MOLDED FLAT CAPACITOR TUBULAR CAPACITOR NORMALLY STAMPED FOR VALUE COMMERCIAL CODE r2NDjSIGNIFICANTFIGURE MULTIPLIER PrI LI TOLERANCE JAN CODE CAPACITOR SILVER 1571 -MULTIPLIER SIG. VOLTAGE FIG. A 2 -DIGIT VOLTAGE RATING INDICATES 900 V. ADD 2 ZEROS TO END OF 2 DIGIT 1ST l SIGNIFICANT 2ND1 FIG. BOO I WI 1ST I -WORKING VOLTS BLACK MORE THAN 2NDJ SIGNIFICANT 1ST J FIGURE MULTIPLIER TOLERANCE CHARACTERISTIC NUMBER. Figure 2 STANDARD COLOR CODE FOR RESISTORS AND CAPACITORS The standard color code provides the necessary information required to properly identify color coded resistors and capacitors. Refer to the color code for numerical values and the number of zeros (or multi- plier) assigned to the colors used. A fourth color band on resistors determines the tolerance rating as follows: Gold= 5%, silver -10%. Absence of the fourth band indicates a 20% tolerance rating. Tolerance rating of capacitors is determined by the color code. For example: Red --2%, green =5%, etc. The voltage rating of capacitors is obtained by multiplying the color value by 100. For example: Orange =3 x100, or 300 volts. www.americanradiohistory.com CHAPTER TWENTY -SEVEN Receivers Transceivers and Receiver construction has just about become lost art. Excellent general coverage receivers are available on the market in many price ranges. However, even the most modest of these receivers is relatively expensive, and most of the receivers are designed as a compromise -they must suit the majority of users, and they must be designed with an eye to the price. It is a tribute to the receiver manufacturers that they have done as well as they have. Even so, the c -w man must often pay for a high fidelity audio system and S -meter he never uses, and the phone man must pay for the c -w man's crystal filter. For one amateur, the receiver has too much bandspread; for the next, too little. For economy's sake and for ease a of alignment, low -Q coils are often found in the r -f circuits of commercial receivers, making the set a victim of cross -talk and overloading from strong local signals. Rarely does the purchaser of a commercial receiver realize that he could achieve the results he desires in a home -built receiver if he left off the frills and trivia which he does not need but which he must pay for when he buys a commercial product. The ardent experimenter, however, needs no such arguments. He builds his receiver merely for the love of the game, and the thrill of using a product of his own creation. It is hoped that the receiving equipment to be described in this chapter will awaken the FIGURE 1 COMPONENT NOMENCLATURE CAPACITORS: 1- RESISTORS 1- RESISTANCE VALUES ARE STATED IN OHMS, THOUSANDS VALUES BELOW 99011JFD ARE INDICATED IN UNITS. EXAMPLE. 1SOJJmFD DESIGNATED AS IS0. 2 - VALUES ABOVE 9991J.UFD ARE INDICATED IN DECIMALS. AS .00S. EXAMPLE: OOSJJFD DESIGNATED 3- 4- (K), AND MEGOHMS (MI. EXAMPLE, 270 OHMS = 270 4700 OHMS = 4.7 H 33,000 OHMS = 33 H 100.000 OHMS = 100 N OR 0.1 33.000,000 OHMS' 33 M OF OHMS OTHER CAPACITOR VALUES ARE AS STATED. EXAMPLE: ,01JFD, 0.51J1JFD, ETC. M 2- ALL RESISTORS ARE 1 -WATT COMPOSITION TYPE UNLESS OTHERWISE NOTED. WATTAGE NOTATION IS THEN INDICATED BELOW RESISTANCE VALUE. TYPE OF CAPACITOR IS INDICATED BENEATH THE VALUE DESIGNATION. SM = SILVER MICA C = CERAMIC EXAMPLE: M' MICA 47 N 0.5 P' PAPER EXAMPLE. 250 EXAMPLE' 6- M .001 INDUCTORS. ELECTROLYTIC OR "FILTER INDICATED BELOW CAPACITY DESIGNATION. 5- VOLTAGE RATING CAPACITOR IS .01 P C MICROHENRIES= JAI OF 20 10 450 ' 600. MILLIHENRIES= HENRIES= 25 10 THE CURVED LINE IN CAPACITOR SYMBOL REPRESENTS THE OUTSIDE FOIL 'GROUND OF PAPER CAPACITORS. THE NEGATIVE ELECTRODE OF ELECTROLYTIC CAPACITORS, OR THE ROTOR OF VARIABLE CAPACITORS. tII II MH H SCHEMATIC SYMBOLSI OR - -LT IL CONDUCTORS JOINED CONDUCTORS CROSSING BUT NOT JOINED 526 www.americanradiohistory.com CHASSIS GROUNO Noise HANDBOOK Miscellaneous There are several other potential noise sources on a passenger vehicle, but they do not necessarily give trouble and therefore require attention only in some cases. The heat, oil pressure, and gas gauges can cause a rasping or scraping noise. The gas gauge is the most likely offender. It will cause trouble only when the car is rocked or is in motion. The gauge units and panel indicators should both be by- passed with the 0.1 -µfd. paper and 0.00l-pfd. mica or ceramic combination previously described. At high car speeds under certain atmospheric conditions corona static may be encountered unless means are taken to prevent it. The receiving-type auto whips which employ a plastic ball tip are so provided in order to minimize this type of noise, which is simply a discharge of the frictional static built up on the car. A whip which ends in a relatively sharp metal point makes an ideal discharge point for the static charge, and will cause corona trouble at a much lower voltage than if the tip were hooded with insulation. A piece of Vinylite sleeving slipped over the top portion of the whip and wrapped tightly with heavy thread will prevent this type of static discharge under practically all conditions. An alternative arrangement is to wrap the top portion of the whip with Scotch brand electrical tape. Generally speaking it is undesirable from the standpoint of engine performance to use both spark -plug suppressors and a distributor suppressor. Unless the distributor rotor clearance is excessive, noise caused by sparking of the distributor rotor will not be so bad but what it can be handled satisfactorily by a noise limiter. If not, it is preferable to shield the hot lead between ignition coil and distributor rather than use a distributor suppressor. Suppression 525 In many cases the control rods, speedometer cable, etc., will pick up high- tension noise under the hood and conduct it up under the dash where it causes trouble. If so, all control rods and cables should be bonded to the fire wall (bulkhead) where they pass through, using a short piece of heavy flexible braid of the type used for shielding. In some cases it may be necessary to bond the engine to the frame at each rubber engine mount in a similar manner. If a rear mountéd whip is employed the exhaust tail pipe also should be bonded to the frame if supported by rubber mounts. Determining the source of certain types of noise is made difficult when several things are contributing to the noise, because elimination of one source often will make little or no apparent difference in the total noise. The following procedure will help to isolate and identify various types of noise. Ignition noise will be present only when the ignition is on, even though the engine is turning over. Generator noise will be present when the motor is turning over, regardless of whether the ignition switch is on. Slipping the drive belt off will kill it. Gauge noise usually will be present only when the ignition switch is on or in the "left" position provided on some cars. Wheel static when present will persist when the car clutch is disengaged and the ignition switch turned off (or to the left position), with the car coasting. Body noise will be noticeably worse on a bumpy road than on a smooth road, particularly at low speeds. Locating Noise Sources www.americanradiohistory.com 524 Mobile THE Equipment of the measures may already have been taken when the auto receiver was installed. First either install a spark plug suppressor on each plug, or else substitute Autolite resistor plugs. The latter are more effective than suppressors, and on some cars ignition noise is reduced to a satisfactory level simply by installing them. However, they may not do an adequate job alone after they have been in use for a while, and it is a good idea to take the following additional measures. Check all high tension connections for gaps, particularly the "pinch fit" terminal connectors widely used. Replace old high tension wiring that may have become leaky. Check to see if any of the high tension wiring is cabled with low tension wiring, or run in the same conduit. If so, reroute the low tension wiring to provide as much separation as practicable. By-pass to ground the 6-volt wire from the ignition coil to the ignition switch at each end with a 0.1-pfd. molded case paper capacitor in parallel with a .001 -µfd. mica or ceramic, using the shortest possible leads. Check to see that the hood makes a good ground contact to the car body at several points. Special grounding contactors are available for attachment to the hood lacings on cars that otherwise would present a grounding problem. If the high -tension coil is mounted on the dash, it may be necessary to shield the high tension wire as far as the bulkhead, unless it already is shielded with armored conduit. static is either static by rotation of the tires and brake drums, or is noise generated by poor contact between the front wheels and the axles (due to the grease in the bearings). The latter type of noise seldom is caused by the rear wheels, but tire static may of course be generated by all four tires. Wheel static can be eliminated by insertion of grounding springs under the front hub caps, and by inserting "tire powder" in all inner tubes. Both items are available at radio parts stores and from most auto radio dealers. Wheel Static Wheel electricity generated Voltage Regulator Hash Certain voltage regulators generate an objectionable amount of "hash" at the higher frequencies, particularly in the v -h-f range. A large by-pass will affect the operation of the regulator and possibly damage the points. A small by -pass can be used, however, without causing trouble. At frequencies above the frequency at which the hash becomes objectionable (approximately 20 Mc. or so) a small by -pass is quite effective. A 0.001 -µfd. RADI O mica capacitor placed from the field terminal of the regulator to ground with the shortest possible leads often will produce sufficient improvement. If not, a choke consisting of about 60 turns of no. 18 d.c.c. or bell wire wound on a h -inch form can be added. This should be placed right at the regulator terminal, and the 0.001-µfd. by -pass placed from the generator side of the choke to ground. "whine" often can be satisfactorily suppressed from 550 kc. to 148 Mc. simply by by- passing the armature terminal to ground with a special Generator Whine Generator "auto radio" by-pass of 0.25 or 0.5 pfd. in parallel with a 0.001 -µfd. mica or ceramic capacitor. The former usually is placed on the generator when an auto radio is installed, but must be augmented by a mica or ceramic capacitor with short leads in order to be effective at the higher frequencies as well as on the broadcast band. When more drastic measures are required, special filters can be obtained which are designed for the purpose. These are recommended for stubborn cases when a wide frequency range is involved. For reception only over a comparatively narrow band of frequencies. such as the 10 -meter amateur band, a highly effective filter can be improvised by connecting between the previously described parallel by -pass capacitors and the generator armature terminal a resonant choke. This may consist of no. 10 enamelled wire wound on a suitable form and shunted with an adjustable trimmer capacitor to permit resonating the combination to the center of the frequency band involved. For the 10 -meter band 11 turns close wound on a one -inch form and shunted by a 3-30 µµfd. compression -type mica trimmer is suitable. The trimmer should be adjusted experimentally at the center frequency. When generator whine shows up after once being satisfactorily suppressed, the condition of the brushes and commutator should be checked. Unless a by -pass capacitor has opened up, excessive whine usually indicates that the brushes or commutator are in need of attention in order to prevent damage to the generator. Loose linkages or body or frame joints anywhere in the car are potential static producers when the car is in motion, particularly over a rough road. Locating the source of such noise is difficult, and the simplest procedure is to give the car a thorough tightening up in the hope that the offending poor contacts will be caught by the procedure. The use of braided bonding straps between the various sections of the body of the car also may prove helpful. Body Static www.americanradiohistory.com HANDBOOK Noise Suppression 523 When using a PE -103A, or any dynamotor for that matter, it may be necessary to devote 660 PE-103A one set of contacts on one of the control relays to breaking the plate or screen voltage to the transmitter oscillator, If these are supplied by the dynamotor, because the output of a dynamotor takes a moment to fall to zero when the primary power is removed. 6 V. INPUT 300 60 400 100 150 200 250 300 OUTPUT CURRENT, MA. Vehicular Noise 26-5 Suppression Figure 11 APPROXIMATE OUTPUT VOLTAGE LOAD CURRENT FOR A PE -103A DYNAMOTOR VS. less the 12 -volt brushes will last almost as long as the 6-volt brushes. The reason that these particular dynamo tors can be operated in this fashion is that there At 150 ma. or are two 6-volt windings on the armature, and for 12 -volt operation the two are used in series with both commutators working. The arrangement described above simply substitutes for the regular 6-volt winding the winding and commutator which ordinarily came into operation only on 12 -volt operation. Some operators have reported that the regulation of the PE103A may be improved by operating both commutators in parallel with the 6-volt line. The three wires now coming out of the dynamotor are identified as follows: The smaller wire is the positive high voltage. The heavy wire leaving the same grommet is positive 6 volts and negative high voltage. The single heavy wire leaving the other grommet is negative 6 volts. Whether the car is positive or negative ground, negative high voltage can be taken as car -frame ground. With the negative of the car battery grounded, the plate current can return through the car battery and the armature winding. This simply puts the 6 volts in series with the 500 volts and gives 6 extra volts plate voltage. The trunk of a car gets very warm in summer, and if the transmitter and dynamotor are mounted in the trunk it is recommended that the end housings be left off the dynamotor to facilitate cooling. This is especially important in hot climates if the dynamotor is to be loaded to more than 200 ma. When replacing brushes on a PE -103A check to see if the brushes are marked negative and positive. If so, be sure to install them accordingly, because they are not of the same material. The dynamotor will be marked to show which holder is negative. Satisfactory reception on frequencies above the broadcast band usually requires greater attention to noise suppression measures. The required measures vary with the particular vehicle and the frequency range involved. Most of the various types of noise that may be present in a vehicle may be broken down into the following main categories: (1) Ignition noise. (2) Wheel static (tire static, brake static, and intermittent ground via front wheel bearings). (3) tacts. "Hash" from voltage regulator con- (4) "Whine" from generator commutator segment make and break. (5) Static from scraping connections between various parts of the car. There is no need to suppress ignition noise completely, because at the higher frequencies ignition noise from passing vehicles makes the use of a noise limiter mandatory anyway. However, the limiter should not be given too much work to do, because at high engine speeds a noisy ignition system will tend to mask weak signals, even though with the limiter working, ignition "pops" may appear to be completely eliminated. Another reason for good ignition suppression at the source is that strong ignition pulses contain enough energy when integrated to block the a-v -c circuit of the receiver, causing the gain to drop whenever the engine is speeded up. Since the a-v -c circuits of the receiver obtain no benefit from a noise dipper, it is important that ignition noise be suppressed enough at the source that the a-v -c circuits will not be affected even when the engine is running at high speed. The following procedure should be found adequate for reducing the ignition noise of practically any passenger car to a level which the dipper can handle satisfactorily at any engine speed at any frequency from 500 kc. to 148 Mc. Some Ignition Noise www.americanradiohistory.com 522 Mobile THE Equipment RING TIP 11 PRESS -TO SWITCH SHELL (GROUND) fYI MIRE PLUG OF -TALK FigUrr IO STANDARD CONNECTIONS FOR THE PUSH -TO -TALK SWITCH ON A HAND. HELD SINGLE- BUTTON CARBON MICROPHONE RADI O a time, and the average "on" time should not be more than half the average "off" time. The output voltage vs. current drain is shown approximately in figure 11. The exact voltage will depend somewhat upon the loss resistance of the primary connecting cable and whether or not the battery is on charge. The primary current drain of the dynamotor proper (excluding relays) is approximately 16 amperes at 100 ma., 21 amperes at 160 ma., 26 amperes at 200 ma., and 31 amperes at 250 ma. microphones on the surplus market use these connections. There is an increasing tendency among mobile operators toward the use of microphones having better frequency and distortion characteristics than the standard single -button type. The high- impedance dynamic type is probably the most popular, with the ceramic crystal type next in popularity. The conventional crystal type is not suitable for mobile use since the crystal unit will be destroyed by the high temperatures which can be reached in a closed car parked in the sun in the summer time. The use of low -level microphones in mobile service requires careful attention to the elimination of common -ground circuits in the microphone lead. The ground connection for the shielded cable which runs from the transmitter to the microphone should be made at only one point, preferably directly adjacent to the grid of the first tube in the speech amplifier. The use of a low-level microphone usually will require the addition of two speech stages (a pentode and a triode), but these stages will take only a milliampere or two of plate current, and 150 ma. per tube of heater current. Because of its availability on the surplus market at a low price and its suitability for use with about as powerful a mobile transmitter as can be employed in a passenger car without resorting to auxiliary batteries or a special generator, the PE -103A is probably the most widely used dynamotor for amateur work. Therefore some useful information will be given on this unit. The nominal rating of the unit is 500 volts and 160 ma., but the output voltage will of course vary with load and is slightly higher with the generator charging. Actually the 160 ma. rating is conservative, and about 275 ma. can be drawn intermittently without overheating, and without damage or excessive brush or commutator wear. At this current the unit should not be run for more than 10 minutes at PE -103A Dynemotor Power Unit Only a few of the components in the base are absolutely necessary in an amateur mobile installation, and some of them can just as well be made an integral part of the transmitter if desired. The base can be removed for salvage components and hardware, or the dynamotor may be purchased without base. To remove the base proceed as follows: Loosen the four thumb screws on the base plate and remove the cover. Remove the four screws holding the dynamotor to the base plate. Trace the four wires coming out of the dynamotor to their terminals and free the lugs. Then these four wires can be pulled through the two rubber grommets in the base plate when the dynamotoris separated from the base plate. It may be necessary to bend the eyelets in the large lugs in order to force them through the gromm et s. Next remove the two end housings on the dynamotor. Each is held with two screws. The high -voltage commutator is easily identified by its narrower segments and larger diameter. Next to it is the 12 -volt commutator. The 6volt commutator is at the other end of the armature. The 12 -volt brushes should be removed when only 6 -volt operation is planned, in order to reduce the drag. If the dynamotor portion of the PE-103A power unit is a Pioneer type VS-25 or a Russell type 530- (most of them are), the wires to the 12 -volt brush holder terminals can be cross connected to the 6-volt brush holder terminals with heavy jumper wires. One of the wires disconnected from the 12-volt brush terminals is the primary 12-volt pigtail and will come free. The other wire should be connected to the opposite terminal to form one of the jumpers. With this arrangement it is necessary only to remove the 6-volt brushes and replace the 12 -volt brushes in case the 6-volt commutator becomes excessively dirty or worn or starts throwing solder. No difference in output voltage will be noted, but as the 12 -volt brushes are not as heavy as the 6-volt brushes it is not permissible vi draw more than about 150 ma. except for emergency use until the 6-volt commutator can be turned down or repaired. www.americanradiohistory.com HANDBOOK Control Do not attempt to control too many relays, particularly heavy duty relays with large coils, by means of an ordinary push -to-talk switch on a microphone. These contacts are not designed for heavy work, and the inductive kick will cause more sparking than the contacts on the microphone switch are designed to handle. It is better to actuate a single relay with the push -to-talk switch and then control all other relays, including the heavy duty contactor for the dynamotor or vibrator pack, with this relay. The procedure of operating only one relay directly by the push -to-talk switch, with all other relays being controlled by this control relay, will eliminate the often -encountered difficulty where the shutting down of one item of equipment will close relays in other items as a result of the coils of relays being placed in series with each other and with heater circuits. A recommended general control circuit, where one side of the main control relay is connected to the hot 6 -volt circuit, but all other relays have one side connected to ground, is illustrated in figure 9. An additional advantage of such a circuit is that only one control wire need be run to the coil of each additional relay, the other side of the relay coils being grounded. The heavy -duty 6 -volt solenoid -type contactor relays such as provided on the PE -103A and used for automobile starter relays usually draw from 1.5 to 2 amperes. While somewhat more expensive, heavy -duty 6-volt relays of conventional design, capable of breaking 30 amperes at 6 volts d.c., are available with coils drawing less than 0.5 ampere. When purchasing relays keep in mind that the current rating of the contacts is not a fixed value, but depends upon (1) the voltage, (2) whether it is a.c. or d.c., and (3) whether the circuit is purely resistive or is inductive. If in doubt, refer to the manufacturer's recommendations. Also keep in mind that a dynamotor presents almost a dead short until the armature starts turning, and the starting relay should be rated at considerably more than the normal dynamotor current. The most generally used microphone for mobile work is the single -button carbon. With a high -output-type microphone and a high-ratio microphone transformer, it is possible when "close talking" to drive even a pair of push pull 6L6's without resorting to a speech amplifier. However, there is a wide difference in the output of the various type single button microphones, and a wide difference in the amount of step up obtained with different type microphone transformers. So at least one Microphones and Circuits speech stage usually is desirable. One of the most satisfactory single button PUSH- TO-TALK SWITCH ON MIKE r PUSH -TO RELAY Circuits 521 -TALK Ll I V rY ALTERNATE CONTROL SWITCH MAIN POWER RECEIVER MUTING RELAY RELAY ANTENNA CHANGEOVER RELAY ANY OTHER RELAYS Figure 9 RELAY CONTROL CIRCUIT Simplified schematic of the recommended relay control circuit for mobile transmitters. The relatively small push -to -talk relay is controlled by the button on the microphone or the communications switch. Then one of the contacts on this relay controls the other relays of t he transmitter; one side of the coil of all the additional relays controlled should be grounded. microphones is the standard Western Electric type F -1 unit (or Automatic Electric Co. equivalent). This microphone has very high output when operated at 6 volts, and good fidelity on speech. When used without a speech amplifier stage the microphone transformer should have a 50 -ohm primary (rather than 200 or 500 ohms) and a secondary of at least 150,000 ohms and preferably 250,000 ohms. The widely available surplus type T-17 microphone has higher resistance (200 to 500 ohmsl and lower output, and usually will require a stage of speech amplification except when used with a very low power modulator stage. Unless an F -1 unit is used in a standard housing, making contact to the button presents somewhat of a problem. No serious damage will result from soldering to the button if the connection is made to one edge and the soldering is done very rapidly with but a small amount of solder, so as to avoid heating the whole button. A sound-powered type microphone removed from one of the chest sets available in the surplus market will deliver almost as much voltage to the grid of a modulator stage when used with a high -ratio microphone transformer as will an F -1 unit, and has the advantage of not requiring button current or a "hash filter." This is simply a dynamic microphone designed for high output rather than maximum fidelity. The standardized connections for a single button carbon microphone provided with push to -talk switch are shown in figure 10. Practically all hand-held military-type single -button www.americanradiohistory.com THE Mobile Equipment 520 RADIO Construction and 26 -4 Installation of Mobile Equipment Figure PI- 8 NETWORK ANTENNA COUPLER The pi- network antenna coupler is particularly satisfactory for mobile work since the coupler affords some degree of harmonic reduction, provides o coupling variation to meet varying load conditions caused by frequency changes, and can cancel out reactance presented to the transmitter at the end of the antenna transmission line. For use of the coupler on the 3.9 -Mc. band CI should hove a maximum capacitance of about 250 µµfd., LI should be about 9 microhenrys (30 turns 1" dia. by 2" long), and C2 may include a fixed and a variable element with maximum capacitance of about 1400 µµ/d. A 100 -µµfd. variable capacitor will be suitable at C1 for the 14 -Mc. and 28 -Mc. bonds, with o 350 -44fd. variable at C2. Inductor LI should have an inductance of about 2 microhenrys (11 turns 1" dia. by 1" long) for the 14 -Mc. band, and about 0.8 mlcrohenry (6 turns 1" dia. by 1" long) for the 28 -Mc. band. quency. Or, if the tapped type of coil is used, taps are changed until the proper number of turns for the desired operating frequency is found. This procedure is repeated for the different bands of operation. Feeding the Center- Loaded Antenna After much experimenting it has been found that the most satisfactory method for feeding the coaxial line to the base of a center- loaded antenna is with the pi- network coupler. Figure 8 shows the basic arrangement, with recommended circuit con- stants. It will be noted that relatively large values of capacitance are required for all bands of operation, with values which seem particularly large for the 75-meter band. But reference to the discussion of pi- network tank circuits in Chapter 13 will show that the values suggested are normal for the values of impedance, impedance transformation, and operating Q which are encountered in a mobile installation of the usual type. It is recommended that the following measures be taken when constructing mobile equipment, either transmitting or receiving, to ensure trouble -free operation over long periods: Use only a stiff, heavy chassis unless the chassis is quite small. Use lock washers or lock nuts when mounting components by means of screws. Use stranded hook -up wire except where r-f considerations make it inadvisable (such as for instance the plate tank circuit leads in a amplifier). Lace and tie leads wherever necessary to keep them from vibrating or flopping around. Unless provided with gear drive, tuning capacitors in the large sizes will require a rotor lock. Filamentary (quick heating) tubes should be mounted only in a vertical position. The larger size carbon resistors and mica capacitors should not be supported from tube socket pins, particularly from miniature sockets. Use tie points and keep the resistor and capacitor "pigtails" short. Generally speaking, rubber shock mounts are unnecessary or even undesirable with passenger car installations, or at least with full size passenger cars. The springing is sufficiently "sott" that well constructed radio equipment can be bolted directly to the vehicle without damage from shock or vibration. Unless shock mounting is properly engineered as to the stiffness and placement of the shock mounts, mechanical- resonance "amplification" effects may actually cause the equipment to v -h -f be shaken more than if the equipment were bolted directly to the vehicle. Surplus military equipment provided with shock or vibration mounts was intended for use in aircraft, jeeps, tanks, gun- firing Naval craft, small boats, and similar vehicles and craft subject to severe shock and vibration. Also, the shock mounting of such equipment is very carefully engineered in order to avoid harmful resonances. To facilitate servicing of mobile equipment, all interconnecting cables between units should be provided with separable connectors on at least one end. The send -receive control circuits of a mobile installation dictated by the design of the equipment, therefore will he left to the ingenuity of reader. However, a few generalizations suggestions are in order. Control Circuits are and the and www.americanradiohistory.com HANDBOOK Mobile A more effective radiator and a better line match may be obtained by making the whip approximately 10% feet long and feeding it with 75 -ohm coax (such as RG-11 /U) via a series capacitor, as shown in figure 6. The relay and series capacitor are mounted inside the trunk, as close to the antenna feedthrou h or base-mount insulator as possible. The 10foot length applies to the overall length from the tip of the whip to the point where the lead in passes through the car body. The leads inside the car (connecting the coaxial cable, relay, series capacitor and antenna lead) should be as short as possible. The outer conductor of both coaxial cables should be grounded to the car body at the relay end with short, heavy conductors. 100 -µµtd. midget variable capacitor is suitable for C1. The optimum setting should A be determined experimentally at the center of the band. This setting then will be satisfactory over the whole band. One suitable coupling arrangement for either a 1/4-wave or 5/16 -wave whip on 10 meters is to use a conventional tank circuit, inductively coupled to a "variable link" coupling loop which feeds the coaxial line. Alternatively, a pi- network output circuit may be used. If the input impedance of the line is very low and the tank circuit has a low C/L ratio, it may be necessary to resonate the coupling loop with series capacitance in order to obtain sufficient coupling. This condition often is encountered with a %-wave whip when the line length approximates an electrical half wave- length. If an all -band center -loaded mobile antenna is used, the loading coil at the center of the antenna may be shorted out for operation of the antenna on the 10 -meter band. The usual type of center-loaded mobile antenna will be between 9 and 11 feet long, including the center- loading inductance which is shorted out. Hence such an antenna may be shortened to an electrical quarter-wave for the 10 -meter band by using a series capacitor as just discussed. Alternatively, if a pi- network is used in the plate circuit of the output stage of the mobile transmitter, any reactance presented at the antenna terminals of the transmitter by the antenna may be tuned out with the pi -network. The All -Bond Center-Loaded Mobile Antenna The great majority of mobile operation on the 14 -Mc. band and below is with center loaded whip antennas. These antennas use an insulated bumper or body mount, with provision for coaxial feed from the base of the antenna to the transmitter, as shown in figure 7. The center -loaded whip antenna must be Antennas 519 CAR BODY UNSHIELDED LOADING COIL RG-56/U LINE TO TRANSMITTER COAXIAL LINE GROUNDED TO FRAME OF CAR ADJACENT TO BASE OF Figure ANTENNA 7 THE CENTER -LOADED WHIP ANTENNA The center -loaded whip antenna, when provided with a topped loading coil or a series of coils, may be used over a wide frequency range. The loading coil may be shorted for use of the antenna on the 10 -meter bond. tuned to obtain optimum operation on the desired frequency of operation. These antennas will operate at maximum efficiency over a range of perhaps 20 kc. on the 75 -meter band, covering a somewhat wider range on the 40meter band, and covering the whole 20 -meter phone band. The procedure for tuning the antennas is discussed in the instruction sheet which is furnished with them, but basically the procedure is as follows: The antenna is installed, fully assembled, with a coaxial lead of RG-58 /U from the base of the antenna to the place where the transmitter is installed. The rear deck of the car should be closed, and the car should be parked in a location as clear as possible of trees, buildings, and overhead power lines. Objects within 15 or 20 feet of the antenna can exert a considerable detuning effect on the antenna system due to its relatively high operating Q. The end of the coaxial cable which will plug into the transmitter is terminated in a link of 3 or 4 turns of wire. This link is then coupled to a grid -dip meter and the resonant frequency of the antenna determined by noting the frequency at which the grid current fluctuates. The coils furnished with the antennas normally are too large for the usual operating frequency, since it is much easier to remove turns than to add them. Turns then are removed, one at a time, until the antenna resonates at the desired frequency. If too many turns have been removed, a length of wire may be spliced on and soldered. Then, with a length of insulating tubing slipped over the soldered joint, turns may be added to lower the resonant fre- www.americanradiohistory.com 518 Mobile Equipment THE 950 RADIO COAX TO RI 7511 COAX TO XuTR Figure 6 RADIATOR FOR 10 METERS If a whip antenna is made slightly longer than one-quarter wave it acts as a slightly better radiator than the usual quarter -wave whip, and it can provide a better match to the antenna transmission line if the reactance is tuned out by a serles capacitor close to the base of the antenna. Capacitor C1 may be a 100 -µµid. midget variable. 5/16 -WAVE WHIP remarks are in order on the subject of feed and coupling systems. Figure 5 A CENTER LOADED 80 -METER WHIP USING AIR WOUND COIL MAY BE USED WITH HIGH POWERED TRANSMITTERS An anti -corona loop is placed at the top of the whip to reduce loss of power and burning of tip of antenna. Number of turns in coil is critical and adjustable, high -Q coil is refommended. Whip may be used over frequency range of about 15 kilo- cycles without retuning. 26 -3 Antennas for Mobile Work The most popular mobile anretna for 10 -meter operation is a rear -mounted whip approximately 8 feet long, fed with coaxial line. This is a highly satisfactory antenna, but a few 10 -Meter Antennas Mobile The feed point resistance of a resonant quarter -wave rear -mounted whip is approximately 20 to 25 ohms. While the standing -wave ratio when using 50 -ohm coaxial line will not be much greater than 2 to 1, it is nevertheless desirable to make the line to the transmitter exactly one quarter wavelength long electrically at the center of the band. This procedure will minimize variations in loading over the band. The physical length of RG -8 /U cable, from antenna base to antenna coupling coil, should be approximately 5 feet 3 inches. The antenna changeover relay preferably should be located either at the antenna end or the transmitter end of the line, but if it is more convenient physically the line may be broken anywhere for insertion of the relay. If the same rear -mounted whip is used for broadcast -band reception, attenuation of broadcast -band signals by the high shunt capacitance of the low impedance feed line can be reduced by locating the changeover relay right at the antenna lead in, and by running 95 -ohm coax (instead of 50 or 75 ohm coax) from the relay to the converter. Ordinarily this will produce negligible effect upon the operation of the .converter, but usually will make a worthwhile improvement in the strength of broadcast band signals. www.americanradiohistory.com One -Tube HANDBOOK auto -set combination has not proven very satisfactory. The primary reason for this is the fact that the relatively sharp i -f channel of the auto set imposes too severe a limitation on the stability of the high- frequency oscillator in the converter. And if a crystal- controlled beating oscillator is used in the converter, only a portion of the band may be covered by tuning the auto set. The most satisfactory arrangement has been found to consist of a separately mounted i.f., audio, and power supply system, with the converter mounted near the steering column. The i -f system should have a bandwidth of 30 to 100 kc. and may have a center frequency of 10.7 Mc. if standard i -f transformers are to be used. The control head may include the 144 Mc. r -f, mixer, and oscillator sections, and sometimes the first i-f stage. Alternatively, the control head may include only the h -f oscillator, with a broadband r -f unit included within the main receiver assembly along with the i.f. and audio system. Commercially manufactured kits and complete units using this general lineup are available. An alternative arrangement is to build a converter, 10.7 -Mc. i -f channel, and second detector unit, and then to operate this unit in conjunction with the auto -set power supply, audio system, and speaker. Such a system makes economical use of space and power drain, and can be switched to provide normal broadcast-band auto reception or reception through a converter for the h -f amateur bands. A recent development has been the VHF transceiver, typified by the Gonset Communicator. Such a unit combines a crystal controlled transmitter and a tunable VHF receiver together with a common audio system and power supply. The complete VHF station may be packaged in a single cabinet. Various forms of VHF transceivers are shown in the construction chapters of this Handbook. 26-2 tion. A Converter 517 total transmitter power drain of about 80 watts from the car battery (6 volts at 13 amperes, or 12 volts at 7 amperes) is about the maximum that can be allowed under these conditions. For maximum power efficiency it is recommended that a vibrator type of supply be used as opposed to a dynamotor supply, since the conversion efficiency of the vibrator unit is high compared to that of the dynamotor. A second school of thought states that the mobile transmitter should be of relatively high power to overcome the poor efficiency of the usual mobile whip antenna. In this case, the mobile power should be drawn from a system that is independent from the electrical system of the automobile. A belt driven high voltage generator is often coupled to the automobile engine in this type of installation. variation of this idea is to employ complete secondary power system in the car capable of providing 115 volts a.c. Shown in figure 4 is a Leece -Neville three phase alternator mounted atop the engine block, and driven with a fan belt. The voltage regulator and selenium rectifier for charging the car battery from the a -c system replace the usual d -c generator. These new items are mounted in the front of the car radiator. The alternator provides a balanced deltaovtput circuit wherein the line voltage is equal to the coil voltage, but the line current is N/3 times the coil current. The coil voltage is a nominal 6- volts, RMS and three 6.3 volt 25 ampere filament transformers may be connected in delta on the primary and secondary windings to step the 6-volts up to three -phase 115-volts. If A a desired, a special 115 -volt, 3 -phase transformer may be wound which will less space than the three filament formers. Since the ripple frequency of phase d-c power supply will be quite single 10 mfd step -up occupy transa three high, a filter capacitor will suffice. Mobile Transmitters As in the case of transmitters for fixed -station operation, there are many schools of thought as to the type of transmitter which is most suitable for mobile operation. One school states that the mobile transmitter should have very low power drain, so that no modification of the electrical system of the automobile will be required, and so that the equipment may be operated without serious regard to discharging the battery when the car is stopped, or overloading the generator when the car is in mo- Figure 4 LEECE -NEVILLE 3 -PHASE ALTERNATOR IS ENGINE DRIVEN BY AUXILIARY FAN BELT. www.americanradiohistory.com 516 Mobile converter end or the set end of the cable between the converter and receiver. This auxiliary trimmer should have a range of about 3 to 50 µµfd., and may be of the inexpensive compression mica type. with the trimmer cut out and the converter turned off (by- passed by the "in -out" switch), peak the regular antenna trimmer on the auto set at about 1400 kc. Then turn on the converter, with the receiver tuned to 1500 kc., switch in the auxiliary trimmer, and peak this trimmer for maximum background noise. The auxiliary trimmer then can be left switched in at all times except when receiving very weak broadcast band signals. Some auto sets, particularly certain General Motors custom receivers, employ a high -Q high impedance input circuit which is very critical as to antenna capacitance. Unless the shunt capacitance of the antenna (including cable) approximates that of the antenna installation for which the set was designed, the antenna trimmer on the auto set cannot be made to hit resonance with the converter cut out. This is particularly true when a long antenna cable is used to reach a whip mounted at the rear of the car. Usually the condition can be corrected by unsoldering the internal connection to the antenna terminal connector on the auto set and inserting in series a 100 -µµfd. mica capacitor. Alternatively an adjustable trimmer covering at least 50 to 150 µµfd. may be substituted for the 100 -µµfd. fixed capacitor. Then the adjustment of this trimmer and that of the regular antenna trimmer can be juggled back and forth until a condition is achieved where the input circuit of the auto set is resonant with the converter either in or out of the circuit. This will provide maximum gain and image rejection under all conditions of use. Reducing Battery Drain of the Receiver THE Equipment When the receiving installa- tion is used frequently, and particularly when the receiver is used with the car parked, it is desirable to keep the battery drain of the receiver -converter installation at an absolute minimum. A substantial reduction in drain can be made in many receivers, without appreciably affecting their performance. The saving of course depends upon the design of the particular receiver and upon how much trouble and expense one is willing to go to. Some receivers normally draw (without the converter connected) as much as 10 amperes. In many cases this can be cut to about 5 amperes by incorporating all practicable modifications. Each of the following modifications is applicable to many auto receivers. If the receiver uses a speaker with a field coil, replace the speaker with an equivalent PM type. RADIO Practically all 0.3- ampere r-f and a -f voltage amplifier tubes have 0.15-ampere equivalents. In many cases it is not even necessary to change the socket wiring. However, when substituting i -f tubes it is recommended that the i -f trimmer adjustments be checked. Generally speaking it is not wise to attempt to substitute for the converter tube or a -f power output tube. If the a-f output tube employs conventional cathode bias, substitute a cathode resistor of twice the value originally employed, or add an identical resistor in series with the one already in the set. This will reduce the B drain of the receiver appreciably without seriously reducing the maximum undistorted output. Because the vibrator power supply is much less than 100 per cent efficient, a saving of one watt of B drain results in a saving of nearly 2 watts of battery drain. This also minimizes the overload on the B supply when the converter is switched in, assuming that the converter uses B voltage from the auto set. If the receiver uses push -pull output and if one is willing to-accept a slight reduction in the maximum volume obtainable without distortion, changing over to a single ended stage is simple if the receiver employs conventional cathode bias. Just pull out one tube, double the value of cathode bias resistance, and add a 25 -{ád. by -pass capacitor across the cathode resistor if not already by- passed. In some cases it may be possible to remove a phase inverter tube along with one of the a -f output tubes. If the receiver uses a motor driven station selector with a control tube (d-c amplifier), usually the tube can be removed without upsetting the operation of the receiver. One then must of course use manual tuning. While the changeover is somewhat expensive, the 0.6 ampere drawn by a 6X4 or 6X5 rectifier can be eliminated by substituting six 115-volt r -m -s 50 -ma. selenium rectifiers (such as Federal type 402D3200). Three in series are substituted for each half of the full -wave rectifier tube. Be sure to observe the correct polarity. The selenium rectifiers also make a good substitution for an OZ4 or OZ4 -GT which is causing hash difficulties when using the converter. Offsetting the total cost of nearly $4.00 is the fact that these rectifiers probably will last for the entire life of the auto set. Before purchasing the rectifiers, make sure that there is room available for mounting them. While these units are small, most of the newer auto sets employ very compact construction. For reception on the 144 amateur band, and those higher in frequency, the simple converterTwo -Meter Reception www.americanradiohistory.com Mc. Mobile Receiver Installation HANDBOOK KEEP THESE LEADS SHORT OR SHIELD THEM. \ VIBRATOR I SPD T. 6v DC TO HOT SIDE OF VOICE COIL WINDING 515 mitter oscillator is killed instantly, thus avoiding trouble from dynamotor "carry over." The efficiency of this arrangement is good because the current drain on the main high voltage supply for the modulated amplifier and modulator plate(s) is reduced by the amount of current borrowed from the receiver. At least 80 ma. can be drawn from practically all auto sets, at least for a short period, without damage. VIA CONTROL RELAY IN XMTR. 6 V Figure 3 METHOD OF ELIMINATING THE BATTERY DRAIN OF THE RECEIVER VIBRATOR PACK DURING TRANSMISSION If the receiver chassis has room for o midget s.p.d.t. relay, the above arrangement not only silences the receiver on transmit but saves several amperes battery drain. If the normally open contact on the relay is connected to the hot side of the voice coil winding as shown in figure 3 (assuming one side of the voice coil is grounded in accordance with usual practice), the receiver will be killed instantly when switching from receive to transmit, in spite of the fact that the power supply filter in the receiver takes a moment to discharge. However, if a "slow start" power supply (such as a dynamotor or a vibrator pack with a large filter) is used with the transmitter, shorting the voice coil probably will not be required. An alternative and high ly recommended procedure is to make use of the receiver B supply on transmit, instead of disabling it. One disadvantage of the popular PE -103A dynamotor is the fact that its 450 -500 volt output is too high for the low power r -f and speech stages of the transmitter. Dropping this voltage to a more suitable value of approximately 250 volts by means of dropping is wasteful of power, besides causing the plate voltage on the oscillator and any buffer stages to vary widely with tuning. By means of a midget 6Using the Receiver Plate Supply On Transmit resistors volt s.p.d.t. relay mounted in the receiver, connected as shown in figure 2, the B supply of the auto set is used to power the oscillator and other low power stages (and possibly screen voltage on the modulator). On transmit the B voltage is removed from the receiver and converter, automatically silencing the receiver. When switching to receive the trans- It will be noted that with the arrangement of figure 2, plate voltage is supplied to the audio output stage at all times. However, when the screen voltage is removed, the plate current drops practically to zero. The 200 -ohm resistor in series with the normally open contact is to prevent excessive sparking when the contacts close. If the relay feeds directly into a filter choke or large capacitor there will be excessive sparking at the contacts. Even with the arrangement shown, there will be considerable sparking at the contacts; but relay contacts can stand such sparking quite a while, even on d.c., before becoming worn or pitted enough to require attention. The 200 -ohm resistor also serves to increase the effectiveness of the .01 -fifd. r -f by -pass capacitor. One other modification of the auto receiver which may or may not be desirable depending upon the circumstances is the addi- Auxiliary Antenna Trimmer tion of an auxiliary antenna trimmer capacitor. If the converter uses an untuned output circuit and the antenna trimmer on the auto set is peaked with the converter cut in, then it is quite likely that the trimmer adjustment will not be optimum for broadcast -band reception when the converter is cut out. For reception of strong broadcast band signals this usually will not be serious, but where reception of weak broadcast signals is desired the loss in gain often cannot be tolerated, especially in view of the fact that the additional length of antenna cable required for the converter installation tends to reduce the strength of broadcast band signals. If the converter has considerable reserve gain, it may be practicable to peak the antenna trimmer on the auto set for broadcast -band reception rather than resonating it to the converter output circuit. But oftentimes this results in insufficient converter gain, excessive image troubles from loud local amateur stations, or both. The difficulty can be circumvented by incorporation of an auxiliary antenna trimmer connected from the "hot" antenna lead on the auto r e c ei v e r to ground, with a switch in series for cutting it in or out. This capacitor and switch can be connected across either the www.americanradiohistory.com Mobile 514 Equipment ( 6V6 THE OPTIONAL Br r0 6AQ5 OR CONVERTER 1 OVC Br TO REST OF SET 6X4 REGULAR R -C FILTER =OLF 30 200 ..UF 2w B+ 200 TO 250 V. TO xMTR -÷7°' I - -- 1 0 HY V. VIA CONTROL RELAY IN XMTR 6 I I I FROM RECEIVER .i T ((}} Br I ""((}} Ti O Br TO LOW POWERI SPEECH STAGES 1 _ L I I XMTR J I Figure 2 USING THE RECEIVER PLATE SUPPLY FOR THE TRANSMITTER This circuit silences the receiver on transmit, and in addition makes it possible to use the receiver plate supply for feeding the ex. citer and speech amplifier stages In the transmitter. is to mount a small receptacle on the receiver cabinet or chassis, making connection via a matching plug. An Amphenol type 77 -26 receptacle is compact enough to fit in a very small space and allows four connections (including ground for the shield braid). The matching plug is a type 70 -26. To avoid the possibility of vibrator hash being fed into the converter via the heater and plate voltage supply leads, it is important that the heater and plate voltages be taken from points well removed from the power supply portion of the auto receiver. If a single -ended audio output stage is employed, a safe place to obtain these voltages is at this tube socket, the high voltage for the converter being taken from the screen. In the case of a push -pull output stage, however, the screens sometimes are fed from the input side of the power supply filter. The ripple at this point, while sufficiently low for a push -pull audio output stage, is not adequate for a converter without additional filtering. If the schematic shows that RADIO the screens of a push -pull stage are connected to the input side instead of the output side of the power supply filter (usually two electrolytics straddling a resistor in an R -C filter), then follow the output of the filter over into the r -f portion of the set and pick it up there at a convenient point, before it goes through any additional series dropping or isolating resistors, as shown in figure 2. The voltage at the output of the filter usually runs from 200 to 250 volts with typical converter drain and the motor not running. This will increase perhaps 10 per cent when the generator is charging. The converter drain will drop the B voltage slightly at the output of the filter, perhaps 15 to 25 volts, but this reduction is not enough to have a noticeable effect upon the operation of the receiver. If the B voltage is higher than desirable or necessary for proper operation of the converter, a 2 -watt carbon resistor of suitable resistance should be inserted in series with the plate voltage lead to the power receptacle. Usually something between 2200 and 4700 ohms will be found about right. Receiver Disabling on Transmit When the battery drain is high on transmit, as is the case when a PE -103A dynamotor is run at maximum rating and other drains such as the transmitter heaters and auto headlights must be considered, it is desirable to disable the vibrator power supply in the receiver during transmissions. The vibrator power supply usually draws several amperes, and as the receiver must be disabled in some manner anyhow during transmissions, opening the 6 -volt supply to the vibrator serves both purposes. It has the further advantage of introducing a slight delay in the receiver recovery, due to the inertia of the power supply filter, thus avoiding the possibility of a feedback "yoop" when switching from transmit to re- ceive. To avoid troubles from vibrator hash, it is best to open the ground lead from the vibrator by means of a midget s.p.d.t. 6 -volt relay and thus isolate the vibrator circuit from the external control and switching circuit wires. The relay is hooked up as shown in figure 3: Standard 8- ampere contacts will be adequate for this application. The relay should be mounted as close to the vibrator as practicable. Ground one of the coil terminals and run a shielded wire from the other coil terminal to one of the power receptacle connections, grounding the shield at both ends. By -pass each end of this wire to ground with .01 pfd., using the shortest possible leads. A lead is run from the corresponding terminal on the mating plug to the control circuits, to be discussed later. www.americanradiohistory.com HANDBOOK Mobile shown. !Multi- position tone controls tied in with the second detector circuit often permit excessive "leak through." Hence it is recommended that the tone control components be completely removed unless they are confined to the grid of the a -f output stage. If removed, the highs can be attenuated any desired amount by connecting a mica capacitor from plate to screen on the output stage. Ordinarily from .005 to .01 tfd. wilt r:ovide a good compromise between fidelity and reduction of background hiss on weak signals. Usually the s wit c h SW will have to be mounted some distance from the noise limiter components. If the leads to the switch are over approximately 1 ii in c h e s long, a piece of shield braid should be slipped over them and grounded. The same applies to the "hot" leads to the volume control if not already shielded. Closing the switch disables the limiter. This may be desirable for reducing distortion on broadcast reception or when checking the intensity of ignition noise to determine the effectiveness of suppression measures taken on the car. The switch also permits one to check the effectiveness of the noise clipper. The 22,000 -ohm decoupling resistor at the bottom end of the i -f transformer secondary is not critical, and if some other value already is incorporated inside the shield can it may be left alone so long as it is not over 47,000 ohms, a common value. Higher values must be replaced with a lower value even if it requires a can opener, because anything over 47,000 ohms will result in excessive loss in gain. There is some loss in a -f gain inherent in this type of limiter anyhow (slightly over 6 db), and it is important to minimize any additional loss. It is important that the total amount of capacitance in the RC decoupling (r -f) filter not exceed about 100 µµfd. With a value much greater than this "pulse stretching" will occur and the effectiveness of the noise clipper will be reduced. Excessive capacitance will reduce the amplitude and increase the duration of the ignition pulses before they reach the clipper. The reduction in pulse amplitude accomplishes no good since the pulses are fed to the clipper anyhow, but the greater duration of the lengthened pulses increases the audibility and the blanking interval associated with each pulse. If a shielded wire to an external dipper is employed, the r-f by -pass on the "low" side of the RC filter may be eliminated since the capacitance of a few feet of shielded wire will accomplish the same result as the by -pass capacitor. The switch SW is connected in such a manner that there is practically no change in gain with the limiter in or out. If the auto set does not have any reserve gain and more gain is Receivers 513 needed on weak broadcast signals, the switch can be connected from the hot side of the volume control to the j unction of the 22,000, 270,000 and 1 megohm resistors instead of as shown. This will provide approximately 6 db more gain when the clipper is switched out. Many late model receivers are provided with an internal r -f gain control in the cathode of the r -f and /or i -f stage. This control should be advanced full on to provide better noise limiter action and make up for the loss in audio gain introduced by the noise clipper. Installation of the noise clipper often detunes the secondary of the last i -f transformer. This should be repeaked before the set is permanently replaced in the car unless the trimmer is accessible with the set mounted in place. Additional clipper circuits will be found in the receiver chapter of this llandbook. Selectivity While not of serious concern on 10 meters, the lack of selectivity exhibited by a typical auto receiver will result in QRM difficulty on 20 and 75 meters. A typical auto set has only two i -f transformers of relatively low -Q design, and the second one is loaded by the diode detector. The skirt selectivity often is so poor that a strong local will depress the a.v.c. when listening to a weak station as much as 15 kc. different in frequency. One solution is to add an outboard i-f stage employing two good quality double -tuned transformers (not the midget variety) connected "back -to- back" through a small coupling capacitance. The amplifier tube (such as a 6BA6) should be biased to the point where the gain of the outboard unit is relatively small (1 or 2), assuming that the receiver already has adequate gain. If additional gain is needed, it may be provided by the outboard unit. Low- capacitance shielded cable should be used to couple into and out of the outboard unit, and the unit itself should be thoroughly shielded. Such an outboard unit will sharpen the nose selectivity slightly and the skirt selectivity greatly. Operation then will be comparable to a home -station communications receiver, though selectivity will not be as good as a receiver employing a 50 -kc. or 85 -kc. "Q5'er." Obtaining Power for the Converter While the set is on the bench for installation of the noise clipper, provision should be made for obtaining filament and plate voltage for the converter, and for the exciter and speech amplifier of the transmitter, if such an arrangement is to be used. To permit removal of either the converter or the auto set from the car without removing the other, a connector should be provided. The best method www.americanradiohistory.com 512 Mobile Equipment THE sary, and it is recommended that a noise clipper be installed without confirming the necessity therefor. It has been found that quiet reception sometimes may be obtained on 75 meters simply by the use of resistor type plugs, but after a few thousand miles these plugs often become less effective and no longer do a fully adequate job. Also, a noise clipper in- RADIO TO A F. POWER AMP. I F TRANS sures against ignition noise from passing trucks and "un- suppressed" cars. On 10 meters a noise clipper is a "must" in any case. There are certain things that should be done to the auto set when it is to be used with a converter, and they might as well be done all at the same time, because "dropping" an auto receiver and getting into the chassis to work on it takes quite a little time. First, however, check the circuit of the auto receiver to see whether it is one of the few receivers which employ circuits which complicate connection of a noise clipper or a converter. If the receiver is yet to be purchased, it is well to investigate these points ahead of Modifying the Auto Receiver time. If the receiver uses a negative B resistor strip for bias (as evidenced by the cathode of the audio output stage being grounded), then the additional plate current drain of the converter will upset the bias voltages on the various stages and probably cause trouble. Because the converter is not on all the time, it is not practical simply to alter the resistance of the bias strip, and major modification of the receiver probably will be required. The best type of receiver for attachment of a converter and noise clipper uses an r -f stage; permeability tuning; single unit construction (except possibly for the speaker); push button tuning rather than a tuning motor; a high vacuum rectifier such as a 6X4 (rather than an OZ4 or a synchronous rectifier); a 6SQ7 (or miniature or Loctal equivalent) with grounded cathode as second detector, first audio, and a.v.c.; power supply negative grounded directly (no common bias strip); a PM speaker (to minimize battery drain); and an internal r -f gain control (indicating plenty of built -in reserve gain which may be called upon if necessary). Many current model auto radios have all of the foregoing features, and numerous models have most of them, something to keep in mind if the set is yet to be purchased. noise limiter either may be built into the set or purchased as a commercially manufactured unit for "outboard" connection via shielded wires. If the receiver employs a 6SQ7 (or Loctal or miniature equivalent) in a conventional circuit, it is a simple matter to build in a noise clipper by Noise Limiters A Figure 1 SERIES -GATE NOISE LIMITER FOR AUTO RECEIVER Auto receivers using a 6SQ7, 786, 7X7, or 6A T6 as second detector and a.v.c. can be converted to the above circuit with but few wiring changes. The circuit hos the advantage of not requiring an additional tube socket for the limiter diode. substituting a 6S8 octal, 7X7 Loctal, or a 6T8 9 -pin miniature as shown in figure 1. When substituting a 6T8 for a 6ÁT6 or similar 7 -pin miniature, the socket must be changed to a miniature type. This requires reaming the socket hole slightly. If the receiver employs cathode bias on the 6SQ7 (or equivalent), and perhaps delayed a.v.c., the circuit usually can be changed to the grounded -cathode circuit of figure 1 without encountering trouble. Some receivers take the r -f excitation for the a -v -c diode from the plate of the i -f stage. In this case, leave the a.v.c. alone and ignore the a -v-c buss connection shown in figure 1 (eliminating the 1- megohm decoupling resistor). If the set uses a separate a -v -c diode which receives r-f excitation via a small capacitor connected to the detector diode, then simply change the circuit to correspond to figure 1. In case anyone might be considering the use of a crystal diode as a noise limiter in conjunction with the tube already in the set, it might be well to point out that crystal diodes perform quite poorly in series -gate noise clippers of the type shown. It will be observed that no tone control is 9 -pin www.americanradiohistory.com CHAPTER TWENTY -SIX Mobile Equipment Design and Installation Mobile operation is permitted on all amateur bands. Tremendous impetus to this phase of the hobby was given by the suitable design of compact mobile equipment. Complete mobile installations may be purchased as packaged units, or the whole mobile station may be home built, according to the whim of the operator. The problems involved in achieving a satisfactory two -way installation vary somewhat with the band, but many of the problems are common to all bands. For instance, ignition noise is more troublesome on 10 meters than on 75 meters, but on the other hand an efficient antenna system is much more easily accomplished on 10 meters than on 75 meters. Also, obtaining a worthwhile amount of trans- mitter output without excessive battery drain is a problem on all bands. 26 -1 Mobile Reception When a broadcast receiver is in the car, the most practical receiving arrangement involves a converter feeding into the auto set. The advantages of good selectivity with good image rejection obtainable from a double conversion superheterodyne are achieved in most cases without excessive "birdie" troubles, a com- difficulty with a double conversion superheterodyne constructed as an integral receiver in one cabinet. However, it is important that the b-c receiver employ an r -f stage in order to provide adequate isolation between the converter and the high frequency oscillator in the b -c receiver. The r -f stage also is desirable from the standpoint of image rejection if the converter does not employ a tuned output circuit (tuned to the frequency of the auto set, usually about 1500 kc.). A few of the late model auto receivers, even in the better makes, do not employ an r -f stage. The usual procedure is to obtain converter plate voltage from the auto receiver. Experience has shown that if the converter does not draw more than about 15 or at most 20 ma. total plate current no damage to the auto set or loss in performance will occur other than a slight reduction in vibrator life. The converter drain can be minimized by avoiding a voltage regulator tube on the converter h -f oscillator. On 10 meters and lower frequencies it is possible to design an oscillator with sufficient stability so that no voltage regulator is required in the converter. With some cars satisfactory 75-meter operation can be obtained without a noise clipper if resistor type spark plugs (such as those made by Autolite) are employed. However, a noise clipper is helpful if not absolutely necesmon 511 www.americanradiohistory.com 510 25 -10 Rotary Beams Indication of Direction The most satisfactory method for indicating the direction of transmission of a rotatable array is that which uses Selsyns or synchros for the transmission of the data from the rotating structure to the indicating pointer at the operating position. A number of synchros and Selsyns of various types are available on the surplus market. Some of them are designed for operation on 115 volts at 60 cycles, some are designed for operation on 60 cycles but at a lowered voltage, and some are designed for operation from 400 -cycle or 800 -cycle energy. This latter type of high- frequency synchro is the most generally available type, and the high- frequency units are smaller and lighter than the 60 -cycle units. Since the indicating synchro must deliver an almost negligible amount of power to the pointer which it drives, the high- frequency types will operate quite satisfactorily from 60-cycle power if the voltage on them is reduced to somewhere between 6.3 and 20 volts. In the case of many of the units available, a connection sheet is provided along with a recommendation in regard to the operating voltage when they are run on 60 cycles. In any event the operating voltage should be held as low as it may be and still give satisfactory transmission of data from the antenna to the operating position. Certainly it should not be necessary to run such a voltage on the units that they become overheated. A suitable Selsyn indicating system is shown in figure 21. Systems using a potentiometer capable of continuous rotation and a milliammeter, along with a battery or other source of direct current, may also be used for the indication of direction. A commercially-available potentiometer (Ohmite RB -2) may be used in conjunction with a 0 -1 d -c milliammeter having a hand calibrated scale for direction indication. 25 -11 traps exert a minimum influence upon the element and it resonates at a frequency determined by the electrical length of the configuration, plus a slight degree of loading contributed by the traps. At some higher frequency (generally about 1.5 times the lowest operating frequency) the outer set of traps are in a parallel resonant condition, placing a high impedance between the element and the tips beyond the traps. Thus, the element resonates at a frequency 1.5 times higher than that determined by the overall length of the element. As the frequency of operation is raised to approximately 2.0 times the lowest operating frequency, the inner set of traps become resonant, effectively disconnecting a larger portion of the element from the driven section. The length of the center section is resonant at the highest frequency of operation. The center section, plus the two adjacent inner sections are resonant at the intermediate frequency of operation, and the complete element is resonant at the lowest frequency of operation. The efficiency of such a system is determined by the accuracy of tuning of both the element sections and the isolating traps. In addition the combined dielectric losses of the traps affect the overall antenna efficiency. As with all multi -purpose devices, some compromise between operating convenience and efficiency must be made with antennas designed to operate over more than one narrow band of frequencies. It is a tribute to the designers of the better multi -band beams that they perform as well as they do, taking into account the theoretical difficulties that must be overcome. r' ISOLATING TRAPS - "Three -Band" Beams popular form of beam antenna introduced during the past few years is the so- called three -band beam. An array of this type is designed to operate on three adjacent amateur bands, such as the ten, fifteen, and twenty meter group. The principle of operation of this form of antenna is to employ parallel tuned circuits placed at critical positions in the elements of the beam which serve to electrically connect and disconnect the outer sections of the elements as the frequency of excitation of the antenna is changed. A typical 'three -band" element is shown in figure 22. At the lowest operating frequency, the tuned it A FEED POINT RESO NANT- T AT NIGNEST FREQUENCY RESONANT 2 AT INTERMEDIATE FREQUENCY RESONANT ; AT LOWEST FREQUENCY Figure 22 TRAP -TYPE "THREE BAND" ELEMENT Isolating traps permit dipole to be self-resonant at three widely different www.americanradiohistory.com frequencies. HANDBOOK -__ Antenna Control Systems ANTENNA ROTATOR CONTROL BOX r S.P D.T. RELAY 509 SOCKET PLUG SOCKET 1 e 1 PLUG R IS-CONTACT JONES PLUGS SYNCHRO. GENERATOR L r D.P.D.T. TOGGLE SWITCH 1 fin SOCKETS ROTARY BEAM CONTROL TO PROP MOTOR INDICATOR J 1 SYNCHRO. PILOT LIGHT ITO 115 V.A.C. i TOGGLE SWITCH SOCKET j PLUG SOCKET L PLUG J DIRECTION INDICATOR Figure 21 SCHEMATIC OF A COMPLETE ANTENNA CONTROL SYSTEM ating position as possible. However, on a particular installation the positions of the current minimums on the transmission line near the transmitter may be checked with the array in the air, and then the array may be lowered to ascertain whether or not the positions of these points have moved. If they have not, and in most cases if the feeder line is strung out back and forth well above ground as the antenna is lowered they will not change, the positions of the last few toward the antenna itself may be determined. Then the calculation of the matching quarter-wave section may be made, the section installed, the standing -wave ratio again checked, and the antenna re- installed in its final location. 25 -9 Antenna Rotation Systems Structures for the rotation of antenna arrays may be divided into two general classes: the rotating mast and the rotating platform. The rotating mast is especially suitable where the transmitting equipment is installed in the garage or some structure away from the main house. Such an installation is shown in figure 19. A very satisfactory rotation mechanism is obtained by the use of a large steering wheel located on the bottom pipe of the rotating mast, with the thrust bearing for the structure located above the roof. If the rotating mast is located a distance from the operating position, a system of pulleys and drive rope may be used to turn the antenna, or a slow speed electric motor may be employed. The rotating platform system is best if a tower or telephone pole is to be used for antenna support. A number of excellent rotating platform devices are available on the market for varying prices. The larger and more expensive rotating devices are suitable for the rdtaof a rather sizeable array for the 14-Mc. band while the smaller structures, such as those designed for rotating a TV antenna are designed for less load and should be used only with a 28 -Mc. or 50 -Mc. array. Most common practice is to install the rotating device atop a platform built at the top of a telephone pole or on the top of a lattice mast of sizeable cross section so that the mast will be self- supporting and capable of withstanding the torque imposed upon it by the rotating platform. A heavy duty TV rotator may be employed for rotation of 6 and 10 -meter arrays. Fifteen and twenty meter arrays should use rotators designed for amateur use such as the Cornell Dubilier HAM -I unit shown in figure 20. www.americanradiohistory.com Rotary 508 T H E B e a m s R A D has been determined previously, and the Antennascope control is turned for a null readingon the meter of the Antennascope. The impedance presented to the Antennascope by the matching device may be read directly on the calibrated dial of the Antennascope. 3. Adjustments should be made to the matching device to present the desired impedance transformation to the Antennascope. If a folded dipole is used as the driven element, the transformation ratio of the dipole must be varied as explained previously in this chapter to provide a more exact match. If a T -match or gamma match system is used, the length of the matching rod may be changed to effect a proper match. If the Antennascope ohmic reading is lower than the desired reading, the length of the matching rod should be increased. If the Antennascope reading is higher than the desired reading, the length of the matching rod should be decreased. After each change in length of the matching rod, the series capacitor in the matching system should be reresonated for best null on the meter of the Antennascope. Raising and Lowering the Array P' 4041,, A practical problem always present when tuning up and matching an array is the physical location of the structure. If the array is atop the mast it is inaccessible for adjustment, and if it is located on stepladders where it can be adjusted easily it cannot be rotated. One encouraging factor in this situation is the fact that experience has shown that if the array is placed 8 or 10 feet above ground on some stepladders for the preliminary tuning process, the raising of the system to its full height will not produce a serious change in the adjustments. So it is usually possible to make preliminary adjustments with the system located slightly greater than head height above ground, and then to raise the antenna to a position where it may be rotated for final adjustments. If the position of the sliding sections as determined near the ground is marked so that the adjustments will not be lost, the array may be raised to rotatable height and the fastening clamps left loose enough so that the elements may be slid in by means of a long bamboo pole. After a series of trials a satisfactory set of lengths can be obtained. But the end results usually come so close to the figures given in figure 5 that a subsequent array is usually cut to the dimensions given and installed as -is. The matching process does not require rotation, but it does require that the antenna proper be located at as nearly its normal oper- Figure 20 HEAVY DUTY ROTATOR SUITABLE FOR AMATEUR BEAMS The new Cornell -Dubilier type HAM -1 rotor has extra heavy motor and gearing system to withstand weight and inertia of amateur array under the buffeting of heavy winds. Steel spur gears and rotor lock prevent "pin wheeling" of antenna. www.americanradiohistory.com I O Tuning the Array HANDBOOK a substantially separate process as just described. Alter the tuning operation is complete, the resonant frequency of the driven element of the antenna should be checked, directly at the center of the driven element if practicable, with a grid dip meter. It is important that the resonant frequency of the antenna be at the center of the frequency band to be covered. If the resonant frequency is found to be much different from the desired frequency, the length of the driven element of the array should be altered until this condition exists. A relatively small change in the length of the driven element will have only a second order effect on the tuning of the parasitic elements of the array. Hence, a moderate change in the length of the driven element may be made without repeating the tuning process for the parasitic elements. When the resonant frequency of the antenna system is correct, the antenna transmission line, with impedance -matching device or network between the line and antenna feed point, is then attached to the array and coupled to a low -power exciter unit or transmitter. Then, preferably, a standing -wave meter is connected in series with the antenna transmission line at a point relatively much more close to the transmitter than to the antenna. However, for best indication there should be 10 to 15 feet of line between the transmitter and the standing -wave meter. If a standing -wave meter is not available the standing -wave ratio may be checked approximately by means of a neon lamp or a short fluorescent tube if twin transmission line is being used, or it may be check- cess of tuning the array is made ed with a thermomilliammeter and a loop, a neon lamp, or an r -f ammeter and a pair of clips spaced a fixed distance for clipping onto one wire of a two -wire open line. If the standing -wave ratio is below 1.5 to 1 Figure 507 it is satisfactory to leave the installation as it is. If the ratio is greater than this range it will be best when twin line or coaxial line is being used, and advisable with open -wire line, to attempt to decrease the s.w.r. It must be remembered that no adjustments made at the transmitter end of the transmission line will alter the SWR on the line. All adjustments to better the SWR must be made at the antenna end of the line and to the device which performs the impedance transformation necessary to match the characteristic impedance of the antenna to that of the transmission line. Before any adjustments to the matching system are made, the resonant frequency of the driven element must be ascertained, as explained previously. If all adjustments to correct impedance mismatch are made at this frequency, the problem of reactance termination of the transmission line is eliminated, greatly simplifying the problem. The following steps should be taken to adjust the impedance transformation: 1. The output impedance of the matching device should be measured. An Antenna scope and a grid -dip oscillator are required for this step. The Antennascope is connected to the output terminals of the matching device. If the driven element is a folded dipole, the Antennascope connects directly to the split section of the dipole. If a gamma match or T -match are used, the Antennascope connects to the transmission -line end of the device. If a Q- section is used, the Antennascope connects to the bottom end of the section. The grid -dip oscillator is coupled to the input terminals of the Antennascope as shown in figure 18. 2. The grid-dip oscillator is tuned to the resonant frequency of the antenna, which 19 ALL -PIPE ROTATING MAST STRUCTURE FOR ROOF INSTALLATION An installation suitable for a building with a pitched roof is shown at (A). At (B) is shown a similar installation for a flat or shed roof. The arrangement as shown is strong enough to support a lightweight 3- element 28 -Mc. array and a light 3- element 50 -Mc. array above the 28 -Mc. array on the end of a 4 -foot length of I/2-inch pipe. The lengths of pipe shown were chosen so that when the system is in the lowered position one can stand on a household ladder and put the beam in position atop the rotating pipe. The lengths may safely be revised upward somewhat if the array is of a particularly lightweight design with low wind resistance. Just before the mast is installed It is a good idea to give the rotating pipe a good smearing of cup grease or waterproof pump grease. To get the lip of the top of the stationary section of 1 %4inch pipe to project above the flange plate, it will be necessary to have a plumbing shop cut a slightly deeper thread inside the flange plate, as well as cutting an unusually long thread on the end of the 1%-inch pipe. It is relatively easy to waterproof this assembly through the roof since the I 1/4-inch pipe is stationary at all times. Be sure to use pipe compound on all the joints and then really tighten these joints with a pair of pipe wrenches. www.americanradiohistory.com 506 Rotary THE Beams RADIO s = JyWyF W ¢ iLZ NW ZZ] O20 JWT FOW N2 Wf iZ óJ WJN ~ < W_ ¢ NN ¢ÓF OV 6f JOS www.americanradiohistory.com O O W U s HANDBOOK Tuning the Array driven onto a wooden dowel, as shown in figure 178. The element may then be mounted upon an aluminum support plate by means of four ceramic insulators. Metal based insulators, such as the Johnson 135 -67 are recommended, since the all- ceramic types may break at the mounting holes when the array is subjected to heavy winds. 25 -8 Tuning the Array Although satisfactory results may be obtained by pre -cutting the antenna array to the dimensions given earlier in this chapter, the occasion might arise when it is desired to make a check on the operation of the antenna before calling the job complete. The process of tuning an array may fairly satisfactorily be divided into two more or less distinct steps: the actual tuning of the array for best front -to-back ratio or for maximum forward gain, and the project of obtaining the best possible impedance match between the antenna transmission line and the feed point of the array. The actual tuning of the array for best front -to -back ratio or maximum forward gain may best be accomplished with the aid of a low -power transmitter feeding a dipole antenna (polarized the same as the array being tuned) at least four or five wavelengths away from the antenna being tuned and located at the same elevation as that of the antenna under test. A calibrated field- strength meter of the remote -indicating type is then coupled to the feed point of the antenna array being tuned. The transmissions from the portable transmitter should be made as short as possible and the call sign of the station making the test should be transmitted at least every ten minutes. It is, of course, possible to tune an array with the receiver connected to it and with a station a mile or two away making transmissions on your request. But this method is more cumbersome and is not likely to give complete satisfaction. It is also possible to carry out the tuning process with the transmitter connected to the array and with the field- strength meter connected to the remote dipole antenna. In this event the indicating instrument of the remote- indicating field - strength meter should be visible from the position where the elements are being tuned. However, when the array is being tuned with the transmitter connected to it there is always the problem of making continual adjustments to the transmitter so that a constant amount of power will be fed to the array under test. Also, if you use this system, Tuning the Array Proper 505 10 watts of power is usually sufficient) and make sure that the antenna transmission line is effectively grounded as far as d -c plate voltage is concerned. The use of the method described in the previous paragraph of course eliminates these problems. One satisfactory method for tuning the array proper, assuming that it is a system with several parasitic elements, is to set the directors to the dimensions given in figure 5 and then to adjust the reflector for maximum forward signal. Then the first director should be varied in length until maximum forward signal is obtained, and so on if additional directors are used. Then the array may be reversed in direction and the reflector adjusted for best front to -back ratio. Subsequent small adjustments may then be made in both the directors and the reflector for best forward signal with a reasonable ratio of front -to -back signal. The adjustments in the directors and the reflector will be found to be interdependent to a certain degree, but if small adjustments are made after the preliminary tuning process a satisfactory set of adjustments for maximum performance will be obtained. It is usually best to make the end sections of the elements smaller in diameter so that they will slip inside the larger tubing sections. The smaller sliding sections may be clamped inside the larger main sections. In making the adjustments described, it is best to have the rectifying element of the remote- indicating field- strength meter directly at the feed point of the array, with a resistor at the feed point of the estimated value of feed -point impedance for the array. use very low power (5 or Matching to the Antenna Transmission Line The problem of matching the impedance of the antenna transmission line to the array is much simplified if the pro- DRIVEN ELEMENT ANTENNASCOK RESONATING CAPACITOR RIDOIR MITER Figure 18 ADJUSTMENT OF GAMMA MATCH BY USE OF ANTENNASCOPE AND GRID -DIP METER www.americanradiohistory.com THE Rotary Beams 504 ELEMENT CLAMP 2 PIECES LINE OF ELEMENTS J- ALUMINUM APPROX. S' - PLATE X LINE BOOM, MADE OF SECTIONS OF STEEL TV MAST OR OF OF ELEMENT ALUMINUM IRRIGATION TUBING 12" - BOOM CLAMP 2 PIECES O PLATE WITH U- BOLTS, REO'O) OR MUFFLER CLAMPS. ELEMENT HELD (2 U- BOLT TO LINE \ SHIM JOINT WITH THIN STRIPS OF ALUMINUM IF NECESSARY h ADJUSTABLE J2' OF BOOM RADIATOR OXEN -YORE CLAMP HOSE CLAMP CENTER S_ECJI ON LADDER SLIT CENTER SECTION TUBE SAT EACH TIP RADIO ADJUSTABLE END. TIP TYPICAL ELEMENT O Figure 16 ELEMENT "PLUMBER'S DELIGHT" ANTENNA ARRAY All-metal con figuration permits rugged, light assembly. Joints are made with U -bolts and metal plates for maximum 3- ELEMENT HELD TO 2%4 BY 2 TV- TYPE U-BOLTS BOLTED TO LADDER BY 2 PIECES OF ANGLE IRON STOCK 2 X rigidity. Figure It is characteristic of the conventional type of multi -element parasitic array such as discussed previously and outlined that the centers of all the elements are at zero r-f potential with respect to ground. It is therefore possible to use a metallic structure without insulators for supporting the various elements of the array. A typical three element array of this type is shown in figure 16. In this particular array, U-bolts and metal plates have been employed to fasten the elements to the boom. The elements are made of telescoping sections of aluminum tubing. The tips of the inner sections of tubing are split, and a tubing clamp is slipped over the joint, as shown in the drawing. Before assembly of the joint, the mating pieces of aluminum are given a thin coat of Penetrox-A compound. (This anti oxidizing paste is manufactured by l3urndy Co., Norwalk, Conn. and is distributed by the General Electric Supply Co.) When the tubes are telesooped and the clamp is tightened, an air -tight seal is produced, reducing corrosion "Plumber's Delight" Construction to a minimum. The boom of the parasitic array may be made from two or three sections of steel TV mast, or it may be made of a single section of aluminum irrigation pipe. This pipe is made by Reynolds Aluminum Co., and others, and may often be purchased via the Sears, Roebuck Co. mail -order department. Three inch pipe may be 17 (A) OXEN -YOKE CLAMP IS DESIGNED FOR ALL METAL ASSEMBLY (B) ALTERNATIVE WOODEN SUPPORTING ARRANGEMENT A wooden ladder may be used to support 70 or 15 meter array. a used for the 10 and 15 meter antennas, and the huskier four inch pipe should be used for a 20 meter beam. Automobile muffler clamps can often be to affix the elements to the support plates. Larger clamps of this type will fasten the plates to the boom. In most cases, the muffler clamps are untreated, and they should be given one or two coats of rust -proof paint to protect them from inclement weather. All bolts, nuts, and washers used in the assembly of the array should be of the plated variety to reduce corrosion and rust. An alternative assembly is to employ the `Oxen Yoke" type of clamps, shown in figure 17. These light- weight aluminum fittings are obtainable from the Continental Electronics and Sound Co., Dayton, 27, Ohio, and are available in a wide range of sizes. If it is desired to use a split driven element for a balanced feed system, it is necessary to insulate the element from the supporting structure of the antenna. The element should be severed at the center, and the two halves used www.americanradiohistory.com Bi- directional HANDBOOK Arrays FLAT -TOP BEAM STUB 503 FOR ROTATABLE ARRAY GAIN OPON -WIRE TO B DB LINE O "TWO OVER TWO OVER TWO* TYPE OF ARRAY Figure GAIN 15 TOTAL NUMBER OF TWO GENERAL TYPES OF BI- DIRECTIONAL ARRAYS Average gain figures are given for both the flat -top beam type of array and for the broadside- eel¡neor array with different numbers of elements. I.B De 5.0 DB ELEMENTS z 7.5 DB 9.0 DB 10.0 DB RAD!AL LOAD 45A BEARING D. FEEDERS N12 WIRE SPACED 10 2' GUY WIRES ROPES TO !NG POSITION A.-THRUST BEARING availability of certain types of constructional materials. But in any event be sure that sound mechanical engineering principles are used in the design of the supporting structure. There are few things quite as discouraging as the picking up of pieces, repairing of the roof, etc., when a newly constructed rotary comes down in the first strong wind. If the principles of mechanical engineering are understood it is wise to calculate the loads and torques which will exist in the various members of the structure with the highest wind velocity which may be expected in the locality of the installation. If this is not possible it will usually be worth the time and effort to look up a friend who understands these principles. Radiating Elements One thing more or less standard about the construction of rotatable antenna arrays is the use of durai tubing for the self- supporting elements. Other materials may be used but an alloy known as 24ST has proven over a period of time to be quite satisfactory. Copper tubing is too heavy for a given strength, and steel tubing, unless copper plated, is likely to add an undesirably large loss resistance to the array. Also, steel tubing, even when plated, is not likely to withstand salt atmosphere such as encountered along the seashore for a satisfactory period of time. Do not use a soft aluminum alloy for the elements unless they will be quite short; 24ST is a hard alloy and is best although there are several other alloys ending in "ST" which will be found to be satisfactory. Do not use an alloy ending in "SO" or "S" in a position in the array where structural strength is important, since these letters designate a metal which has not been heat treated for strength and rigidity. However, these softer alloys, and aluminum electrical conduit, may be used for short radiating elements such as would be used for the 50 -Mc. band or as interconnecting conductors in a stacked array. www.americanradiohistory.com 502 Rotary T Beams H E R A D I O O "LAZY H" WITH REFLECTOR GAIN APPROX. II De BROADSIDE HALF -WAVES WITH REFLECTORS GAIN APPROX. 7 De Figure 14 BROADSIDE ARRAYS WITH PARASITIC REFLECTORS of the arrays illustrated will be greater than the values given due to concentration of the radiated signal at the lower elevation ongles. The apparent gain TWO OVER TWO OVER TWO WITH REFLECTORS GAIN APPROX. II.5 De If six or more elements are used in the type of array shown in figure 15B no matching section will be required between the antenna trans- tuning will apply. However, the factor that a bi- directional array need be rotated through an angle of less than 180° should be considered mission line and the feed point of the antenna. When only four elements are used the antenna is the familiar "lazy H" and a quarter -wave stub should be used for feeding from the antenna transmission line to the feed point of the antenna system. If desired, and if mechanical considerations permit, the gain of the arrays shown in figure 15B may be increased by 3 db by placing a half-wave reflector behind each of the elements at a spacing of one -quarter wave. The array then becomes essentially the same as in this connection. that shown in figure 14C and the same considerations in regard to reflector spacing and 25-7 Construction of Rotatable Arrays A considerable amount of ingenuity may be exercised in the construction of the supporting structure for a rotatable array. Every person has his own ideas as to the best method of construction. Often the most practicable method of construction will be dictated by the HANDBOOK Driven Arrays linear system which will give approximately the same gain as the system of figure 13A, but which requires less boom length and greater total element length. Figure 13C illustrates the familiar lazy -H with driven reflectors (or directors, depending upon the point of view) in a combination which will show wide bandwidth with a considerable amount of forward gain and good front -to -back ratio over the entire frequency coverage. Three practicable types of unidirectional stacked broadside arrays are shown in figure 14. The first type, shown at figure 14A, is the simple "lazy H" type of antenna with parasitic reflectors for each element. (B) shows a simpler antenna array with a pair of folded dipoles spaced one half wave vertically, operating with reflectors. In figure 14C is shown a more complex array with six half waves and six reflectors which will give a very worthwhile amount of gain. In all three of the antenna arrays shown the spacing between the driven elements and the reflectors has been shown as one -quarter wavelength. This has been done to eliminate the requirement for tuning of the reflector, as a result of the fact that a half -wave element spaced exactly one - quarter wave from a driven element will make a unidirectional array when both elements are the same length. Using this procedure will give a gain of 3 db with the reflectors over the gain without the reflectors, with only a moderate decrease in the radiation resistance of the driven element. Actually, the radiation resistance of a half-wave dipole goes down from 73 ohms to 60 ohms when an identical half -wave element is placed one quarter wave behind it. A very slight increase in gain for the entire array (about 1 db) may be obtained at the expense of lowered radiation resistance, the necessity for tuning the reflectors, and decreased bandwidth by placing the reflectors 0.15 wavelength behind the driven elements and making them somewhat longer than the driven elements. The radiation resistance of each element will drop approximately to one -half the value obtained with untunedhalf-wave reflectors spaced one -quarter wave behind the driven elements. Antenna arrays of the type shown in figure 14 require the use of some sort of lattice work for the supporting structure since the arrays occupy appreciable distance in space in all three planes. Unidirectional Stacked Broadside Arrays The requirements for the feed systems for antenna arrays of the type shown in figure 14 are less critical than those for the close- spaced parasitic arrays shown in the previous section. This is a Feed Methods 501 natural result of the fact that a larger number of the radiating elements are directly fed with energy, and of the fact that the effective radiation resistance of each of the driven elements of the array is much higher than the feed-point resistance of a parasitic array. As a consequence of this fact, arrays of the type shown in figure 14 can be expected to cover a somewhat greater frequency band for a specified value of standing -wave ratio than the parasitic type of array. In most cases a simple open -wire line may be coupled to the feed point of the array without any matching system. The standing -wave ratio with such a system of feed will often be less than 2 -to -1. However, if a more accurate match between the antenna transmission line and the array is desired a conventional quarter -wave stub, or a quarter-wave matching transformer of appropriate impedance, may be used to obtain a low standing-wave ratio. 25 -6 Bi- Directional Rotatable Arrays The bi- directional type of array is sometimes used on the 28 -Mc. and 50 -Mc. bands where signals are likely to be coming from only one general direction at a time. Hence the sacrifice of discrimination against signals arriving from the opposite direction is likely to be of little disadvantage. Figure 15 shows two general types of bi- directional arrays. The flattop beam, which has been described in detail earlier, is well adapted to installation atop a rotating structure. When self-supporting elements are used in the flat -top beam the problem of losses due to insulators at the ends of the elements is somewhat reduced. With a single -section flat -top beam a gain of approximately 4 db can be expected, and with two sections a gain of approximately 6 db can be obtained. Another type of bi- directional array which has seen less use than it deserves is shown in figure 15B. This type of antenna system has a relatively broad azimuth or horizontal beam, being capable of receiving signals with little diminution in strength over approximately 40 °, but it has a quite sharp elevation pattern since substantially all radiation is concentrated at the lower angles of radiation if more than a total of four elements is used in the antenna system. Figure 15B gives the approximate gain over a half -wave dipole at the height of the center of the array which can be expected. Also shown in this figure is a type of "rotating mast" structure which is well suited to rotation of this type of array. www.americanradiohistory.com - 500 THE Rotary Beams __Ii/î/ \_----1 10111111---M1111 11---1111 -III M1111 ro 1M ------MM ----III En D 111 11 1111 11NI 11111 Ill 5 1 4 + K 11 IM1111111111111111111111 11111 4 DIRECTIONAL N 1 1....6111 .1111 \A__moimmf1 . B0 110 ó 50 4o u soJ 20 ` 10 Ñ OGAIN ABOUT e De FEED LINE D DIRECTIONAL e SWR Figure RADIO GAIN ABOUT e DB 12 SHORTED STUB LENGTH AND POSITION CHART From the standing wave ratio and current or voltage null position it is possible to determine the theoretically correct length and position of a shorted stub. In actual praca slight discrepancy usually will be found between the theoretical and the experimentally optimized dimensions; therefore it may be necessary to "touch up" the dimensions after using the above data as a FEED LINE tice d4 DIRECTIONAL GAIN ABOUT 10 De starting point. has been decided upon for the stub, and also to determine the SWR. Stub adjustment becomes more critical as the SWR increases, and under conditions of high SWR the current and voltage nulls are more sharply defined than the current and voltage maxima, or loops. Therefore, it is best to locate either a current null or voltage null, depending upon whether a current indicating device or a voltage indicating device is used to check the standing wave pattern. The SWR is determined by means of a "directional coupler," or by noting the ratio of Ema. to Emin or La. to Imin as read on an indicating device. It is assumed that the characteristic impedance of the section of line used as a stub is the same as that of the transmission line proper. It is preferable to have the stub section identical to the line physically as well as electrically. 25 -5 FEED LINE Figure 13 UNIDIRECTIONAL ALL -DRIVEN ARRAYS A unidirectional all -driven end-fire array is shown at (A). (B) shows an array with two half waves in phase with driven reflectors. A Lazy -H array with driven reflectors is shown at (C). Note that the directivity is through the elements with the greatest total feed -line length in arrays such as shown at (B) and (C). Unidirectional Driven Arrays place of a parasitic array of similar dimensions when greater frequency coverage than is available with the yagi type is desired. Figure 13B is a combination end -fire and cobe used in Three types of unidirectional driven arrays are illustrated in figure 13. The array shown in figure 13A is an end -fire system which may www.americanradiohistory.com HANDBOOK The Gamma -FLAT' LINE SWR TO Match 499 RESONANT SECTION 1.O TRANSMITTER SIMPLE OR CONVEX MATCHING STUB Figure 10 THE GAMMA MATCHING SYSTEM See text for details of resonating capacitor piing rings are 10 inches in diameter and are usually constructed of % -inch copper tubing supported one from the rotating structure and one from the fixed structure by means of standoff insulators. The capacitor C in figure 9D is adjusted, after the antenna has been tuned, for minimum standing -wave ratio on the antenna transmission line. The dimensions shown will allow operation with either 14 -Mc. or 28 -Mc. elements, with appropriate adjustment of the capacitor C. The rings must of course be parallel and must lie in a plane normal to the axis of rotation of the rotating structure. The Gamma Match The use of coaxial cable to feed the driven element of a yagi array is becoming increasingly popular. One reason for this increased popularity lies in the fact that the TVI- reduction problem is simplified when coaxial feed line is used from the transmitter to the antenna system. Radiation from the feed line is minimized when coaxial cable is used, since the outer conductor of the line may be grounded at several points throughout its length and since the intense field is entirely confined within the outer conductor of the coaxial cable. Other advantages of coaxial cable as the antenna feed line lie in the fact that coaxial cable may be run within the structure of a building without danger, or the cable may be run underground without disturbing its operation. Also, transmitting -type low -pass filters for 52 ohm impedance are more widely available and are less expensive than equivalent filters for two-wire line. The gamma -match is illustrated in figure 10, and may be looked upon as one -half of a Tmatch. One resonating capacitor is used, placed in series with the gamma rod. The capacitor should have a capacity of 7 µµfd. per meter of wavelength. For 15-meter operation the capacitor should have a maximum capacity of 105 µµfd. The length of the gamma rod determines the impedance transformation between Figure 11 IMPEDANCE MATCHING WITH A CLOSED STUB ON A TWO WIRE TRANSMISSION LINE the transmission line and the driven element of the array, and the gamma capacitor tunes out the inductance of the gamma rod. By adjustment of the length of the gamma rod, and the setting of the gamma capacitor, the SWR on the coaxial line may be brought to a very low value at the chosen operating frequency. The use of an Antennascope, described in the Test Equipment chapter is recommended for precise adjustment of the gamma match. The Matching Stub If an open-wire line is used to feed a low impedance radiator, a section of the transmission line may be employed as a matching stub as shown in figure 11. The matching stub can transform any complex impedance to the characteristic impedance of the transmission line. While it is possible to obtain a perfect match and good performance with either an open stub or a shorted one by observing appropriate dimensions, a shorted stub is much more readily adjusted. Therefore, the following discussion will be confined to the problem of using a closed stub to match a low impedance load to a high impedance transmission line. If the transmission line is so elevated that adjustment of a "fundamental" shorted stub cannot be accomplished easily from the ground, then the stub length may be increased by ex- actly one or two electrical half wavelengths, without appreciably affecting its operation. While the correct position of the shorting bar and the point of attachment of the stub to the line can be determined entirely by experimental methods, the fact that the two adjustments are interdependent or interlocking makes such a cut- and -try procedure a tedious one. Much time can be saved by determining the approximate adjustments required by reference to a chart such as figure 12 and using them as a starter. Usually only a slight "touching up" will produce a perfect match and flat line. In order to utilize figure 12, it is first necessary to locate accurately a voltage node or current node on the line in the vicinity that www.americanradiohistory.com 498 R o t Beams y a r T H E R A D I O OA DIRECT FEED WITH COAXIAL CABLE 52 a. COAXIAL CABLE 0 QUARTER -WAVE TRANSFORMER FEED 75 A TWIN LINE 450 -000 A. LINE Figure © TRANSFORMER MATCHING SYSTEM 20 MC. - 4 TURNS ANT. 14 MC. - METHODS WHERE THE DRIVEN ELEMENT MAY BE BROKEN IN THE CENTER 2" DIA., 2" 1 9 ALTERNATE FEED LONG TURN EACH SIDE TURNS 2" DIA., 2" LONG ANT. TAPPED 2 TURNS EACH SIDE 0 0 COUPLINGLINK ROTARY COILS 10DIAMETER COIL SPACED APPROX. 0.5" C 1 TURN LINKS ARE PARALLEL C IS 200 LUPD VARIABLE 450 -0000. LINE These capacitors should be tuned for minimum SWR on the transmission line. The adjustment of these capacitors should be made at the same time the correct setting of the T -match rods is made as the two adjustments tend to be interlocking. The use of the standing wave meter (described in Test Equipment chapter) is recommended for making these adjustments to the T- match. Four methods of exciting the driven element of a parasitic array are shown in figure 9. The system shown at (A) has proven to be quite satisfactory in the case of an antenna-reflector two element array or in the case of a three-element array with 0.2 to 0.25 wavelength spacing between the elements of the antenna system. The feed -point impedance of the center of the driven element is close enough to the characteristic impedance of the 52 -ohm coaxial cable so that Feed Systems Using Driven Element with Center Feed a the standing -wave ratio on the 52 -ohm coaxial cable is less than 2-to-1.(B) shows an arrangement for feeding an array with a broken driven element from an open -wire line with the aid of a quarter -wave matching transformer. With 465 ohm line from the transmitter to the antenna this system will give a close match to a 12ohm impedance at the center of the driven element. (C) shows an arrangement which uses an untuned transformer with lumped inductance for matching the transmission line to the center impedance of the driven element. Rotary Link Coupling In many cases it is desirable to be able to allow the antenna ar- ray to rotate continuously without regard to snarling of the feed line. If this is to be done some sort of slip rings or rotary joint must be made in the feed line. One relatively simple method of allowing unrestrained rotation of the antenna is to use the method of rotary link coupling shown in figure 9D. The two cou- www.americanradiohistory.com HANDBOOK Matching Systems 497 L frT-u7r. OA L-.j L DELTA MATCH DIMENSIONS SHOWN GIVE APPROX. MATCH TO SOOD AIN - SPACED LINE L Figure 8 AVERAGE DIMENSIONS FOR THE DELTA AND MATCH 1416 L L "T" C -T DI3Dz MATCH n Dn 300 TWIN LINE zoo In many cases it will be desired to use the folded -element or yoke matching system with different sizes of conductors or different spacings than those shown in figure 7. Note, then, that the impedance transformation ratio of these types of matching systems is dependent both upon the ratio of conductor diameters and upon their spacing. The following equation has been given by Roberts (11CA Review, June, 1947) for the determination of the impedance transformation when using different diameters in the two sections of a folded element: Transformation ratio = (1 + \ - Z, z Z, this equation Z, is the characteristic impedance of a line made up of the smaller of the two conductor diameters spaced the center to- center distance of the two conductors in the antenna, and Z, is the characteristic impedance of a line made up of two conductors the size of the larger of the two. This assumes that the feed line will be connected in series with the smaller of the two conductors so that an impedance step up of greater than four will be obtained. If an impedance step up of less than four is desired, the feed line is connected in series with the larger of the two conductors and Z, in the above equation becomes the impedance of a hypothetical line made up of the larger of the two conductors and Z2 is made up of the smaller. The folded v -h -f unipole is an example where the transmission line is connected in series with the a r g e r of the two conductors. In 1 n The conventional 3 -wire match to give an impedance 'multiplication of 9 and the 5 -wire match to give a ratio of approximately 25 are shown in figures 7C and 7D. The 4 -wire match, not shown, will give an impedance transformation ratio of approximately 16. The Delta match and the T -match are shown in figure 8. The delta match has been largely superseded by the newer T- match, however both these systems can be adjusted to give a low value of SWR on 50 to 600 -ohm balanced transmission lines. In the case of the systems shown it will be necessary to make adjustments in the tapping distance along the driven radiator until minimum standing waves on the antenna transmission line are obtained. Since it is sometimes impracticable to eliminate completely the standing waves from the antenna transmission line when using these matching systems, it is common practice to cut the feed line, after standing waves have been reduced to a minimum, to a length which will give satisfactory loading of the transmitter over the desired frequency range of operation. The inherent reactance of the T -match is tuned out by the use of two identical resonating capacitors in series with each leg of the T -rod. These capacitors should each have a maximum capacity of 8 littfd. per meter of wavelength. Thus for 20 meters, each capacitor should have a maximum capacity of at least 160 µµEd. For power up to a kilowatt, 1000 volt spacing of the capacitors is adequate. The Delta Match and T-Match www.americanradiohistory.com 496 Rotary T B ea m s RADIATION Di Di-Da FOR O R.4 D11 FOR MATCH I 5 1.s Rs'8.9 2 s',RrecD R A D Di- 1Dzo. FOR FOLDED -ELEMENT H E D2.z5iegAre=10.5 S1.5 DI1 roa D22.2s1 RAD. 5 D 1- 5 FOR D = *12 WI RE 3- roua D= 1- *12 S 2WIRE FOR 16 =11 - RAD. D= 15 5- 14 18 . w 12 WIR1E roa D. 1S= 1- a0 WIRE POR D. WIRE 12 S =24 I 1 RAD. 32 Figure 7 DATA FOR FOLDED -ELEMENT MATCHING SYSTEMS In all normal applications of 31M12. APPROX. 25 5 -WIRE MATCH the data given the main element as shown is the driven element of a multi -element parasitic array. Directors and reflectors have not been shown for the sake of clarity. R RAD. small strips of polystyrene which have been drilled for both the main element and the small wire and threaded on the main element. The calculation of the operating conditions of folded - element the matching system and the yoke match, as shown in figures 7A and 7B is relatively simple. A selected group of operating. conditions has been shown on the drawing of figure 7. In applying the system it is only necessary to multiply the ratio of feed to radiation resistance (given in the figures to the right of the suggested operating dimensions in figure 7) by the radiation resistance of the antenna system to obtain the impedance of the cable to be used in feeding the array. Approximate values of radiation resistance for a number of commonly used parasitic -element arrays are given The Folded -Element Match Calculations in figure 5. As an example, suppose a 3- element array with 0.15D -0.15R spacing between elements is to be fed by m e an s of a 465 -ohm line constructed of no. 12 wire spaced 2 inches. The approximate radiation resistance of such an antenna array will be 20 ohms. Hence we need a ratio of impedance step up of 23 to obtain a match between the characteristic impedance of the transmission line and the radiation resistance of the driven element of the antenna array. Inspection of the ratios given in figure shows that the fourth set of dimensions given under figure 7B will give a 24 -to -1 step up, which is sufficiently close. So it is merely necessary to use a 1 -inch diameter driven element with a no.8 wire spaced on 1 inch centers (% inch below the outside wall of the 1 -inch tubing) below the 1 -inch element. The no. 8 wire is broken and a 2 -inch insulator placed in the center. The feed line then carries from this insulator down to the transmitter. The center insulator should be supported rigidly from the 1 -inch tube so that the spacing between the piece of tubing and the no. 8 wire will be accurately maintained. 7 www.americanradiohistory.com HANDBOOK H-0.2 A Arrays Stacked Yagi 495 -wi.-13.2 A --4 H 0 2 -+A O. 2 A--+- O. 2 A-.F O 2 7.-.1 501 F Mc DIRECTIONAL DIRECTIONAL A OGAIN ABOUT 12 DB WITH 2 SECTIONS I FEEDER LINE © O GAIN ABOUT IS DB WITH 3 SECTIONS AIN ABOUT 17 DR Figure 6 STACKED YAGI ARRAYS It is possible to attain a relatively large amount of gain over a limited bandwidth with stacked yogi arrays. The two -section array at (A) will give a gain of about 12 db, while adding a third section will bring the gain up to about 15 db. Adding two additional parasitic directors to each section, as at (C) will bring the gain up to about 17 db. higher where the additional section of tubing may be supported below the main radiator element without undue difficulty. The yoke-match is more satisfactory mechanically on the 28- bands since it is only necessary to suspend a wire below the driven element proper. The wire may be spaced below the self-supporting element by means of several Mc. and 14 -Mc. www.americanradiohistory.com 494 TYPE Rotary DRIVEN ELEMENT LENGTH 473 F(MC) 3-ELEMENT 3- ELEMENT THE Beams REFLECTOR LENGTH IST DIRECTOR LENGTH SOI 445 F(MC) .F (MC) F) 741?-41C) 4-ELEMENT 01(1°Z) - -- 2ND DIRECTOR 390 DIRECTOR SPACING BETLENGTH LENGTH WEENELEMENTS -IS .2S -.2S .2 -.2 -.2 S(1SO .1C 5-ELEMENT .IS F(MC) Figure F4(L9C) .2 -.2 -.2-.2 small amount of additional gain may be obtained through use of more than two parasitic elements, at the expense of reduced feed -point impedance and lessened bandwidth. One additional director will add about 1 db, and a second additional director (making a total of five elements including the driven element) will add s l i g ht l y less than one db more. In the v -h -f range, where the additional elements may be added without much difficulty, and where required bandwidths are small, the use of more than two parasitic elements is quite practicable. A Three Elements Parasitic arrays (yagis) may stacked to provide additional gain in the same manner that dipoles may be stacked. Thus if an array of six dipoles would give a gain of 10 db. the substitution of yagi arrays for each of the dipoles would add the gain of one yagi array to the gain obtained with the dipoles. However, the yagi arrays must be more widely spaced than the dipoles to obtain this theoretical improvement. As an example, if six 5- element yagi arrays having a gain of about 10 db were substituted for the dipoles, with appropriate increase in the spacing between the arrays, the gain of the whole system would approach the sum of the two gains, or 20 db. A group of arrays of yagi antennas, with recommended spacing and approximate gains, are illustrated in figure 6. Stacking of Yogi Arrays 25 -4 be Feed Systems for Parasitic (Yogi) Arrays The table of figure 5 gives, in addition to other information, the approximate radiation resistance referred to the center of the driven element of multi- element parasitic arrays. It is obvious, from these low values of radiation APPRO %.RADIATION RESISTANCE 7.5 20 9.S 35 9.S 20 Io.o IS (A I 5 DESIGN CHART FOR PARASITIC ARRAYS (DIMENSIONS GIVEN More Than APPROX. GAIN DO RADIO IN FEET) resistance, that especial care must be taken in materials used and in the construction of the elements of the array to insure that ohmic losses in the conductors will not be an appreciable percentage of the radiation resistance. It is also obvious that some method of iglpedance transformation must be used in many cases to match the low radiation resistance of these antenna arrays to the normal range of characteristic impedance used for antenna transmission lines. A group of possible methods of impedance matching is shown in figures 7, 8, 9 and 10. All these methods have been used but certain of them offer advantages over some of the other methods. Generally speaking it is not mechanically desirable to break the center of the driven element of an array for feeding the system. Breaking the driven element rules out the practicability of building an all -metal or "plumber's delight" type of array, and imposes mechanical limitations with any type of construction. However, when continuous rotation is desired, an arrangement such as shown in figure 9D utilizing a broken driven element with a rotatable transformer for coupling from the antenna transmission line to the driven element has proven to be quite satisfactory. In fact the method shown in figure 9D is probably the most practicable method of feeding the driven element when continuous rotation of the antenna array is required. The feed systems shown in figure 7 will, under normal conditions, show the lowest losses of any type of feed system since the currents flowing in the matching network are the lowest of all the systems commonly used. The "Folded Element" match shown in figure 7A and the "Yoke" match shown in figure 7B are the most satisfactory electrically of all standard feed methods. However, both methods require the extension of an additional conductor out to the end of the driven element as a portion of the matching system. The folded -element match is best on the 50 -Mc. band and www.americanradiohistory.com Parasitic Arrays HANDBOOK 0.2 wavelength between elements becomes possible. Four -element arrays are quite common on the 28 -Mc. and 50 -Mc. bands, and five elements are sometimes used for increased gain and discrimination. As the number of elements is increased the gain and front-to -back ratio increases but the radiation resistance decreases and the bandwidth or frequency range over which the antenna will operate without reduction in effectiveness is decreased. While the elements may consist supported on a wood framework, self-supporting elements of tubing are much to be preferred. The latter type array is easier to construct, looks better, is no more expensive, and avoids the problem of getting sufficiently good insulation at the ends of the elements. The voltages reach such high values towards the ends of the elements that losses will be excessive, unless the insulation is excellent. The elements may be fabricated of thin walled steel conduit, or hard drawn thin -walled copper tubing, but durai tubing is much better. Or, if you prefer, you may purchase tapered copper-plated steel tubing elements designed especially for the purpose. Kits are available complete with rotating mechanism and direction indicator, for those who desire to purchase the whole system ready to put up. Material for Elements of wire The optimum spacing for a two -element array is, as has been mentioned be fore, approximately 0.11 wavelength for a director and 0.13 wavelength for a reflector. However, when both a director and a reflector are combined with the driven element to make up a three-element array the optimum spacing is established by the bandwidth which the antenna will be required to cover. Wide spacing (of the order of 0.25 wavelength between elements) will result in greater bandwidth for a specified maximum standing wave ratio on the antenna transmission line. Smaller spacings may be used when boom length is an important consideration, but for a specified standing-wave ratio and forward gain the frequency coverage will be smaller. Thus the Q of the antenna system will be increased as the spacing between the elements is decreased, resulting in smaller frequency coverage, and at the same time the feed -point impedance of the driven element will be decreased. For broad -band coverage, such as the range from 26.96 to 29.7 Mc. or from 50 to 54 Mc., 0.2 wavelength spacing from the driven element to each of the parasitic elements is recElement Spacing 493 For narrower bandwidth, such as would be adequate for the 14.0 to 14.4 Mc. band or the 144 to 148 Mc. band, the radiator to parasitic element spacing may be reduced to 0.12 wavelength, while still maintaining adequate array bandwidth for the amateur band in question. ommended. Experience has shown that it is practical to cut the prarsitic elements of a three -element parasitic array to a predetermined length before the installation of such an antenna. A pre -tuned antenna such as this will give good signal gain, adequate front -to -back ratio, and good bandwidth factor. By carefully tuning the array after it is in position the gain may be increased by a fraction of a db, and the front -to -back ratio by several db. However the slight improvement in performance is usually not worth the effort expended in tuning time. The closer the lengths of the parasitic elements are to the resonant length of the driven element, the lower will be the feed -point resistance of the driven element, and the smaller will be the bandwidth of the array. Hence, for wide frequency coverage the director should be considerably shorter, and the reflector considerably longer than the driven element. For example, the director should still be less than a resonant half wave at the upper frequency limit of the range wherein the antenna is to be operated, and the reflector should still be long enough to act as a reflector at the lower frequency limit. Another way of stating the same thing is to say, in the case of an array to cover a wide frequency range such as the amateur range from 26.96 to 29.7 Mc. or the width of a low -band TV channel, that the director should be cut for the upper end of the band and the reflector for the lower end of the band. In the case of the 26.96 to 29.7 Mc. range this means that the director should be about 8 per cent shorter than the driven element and the reflector should be about 8 per cent longer. Such an antenna will show a relatively constant gain of about 6 db over its range of coverage, and the pattern will not reverse at any point in the range. Where the frequency range to be covered is somewhat less, such as a high -band TV channel, the 14.0 to 14.4 Mc. amateur band, or the lower half of the amateur 28 -Mc. phone band, the reflector should be about 5 per cent longer than the driven element, and the director about 5 per cent shorter. Such an antenna will perform well over its rated frequency band, will not reverse its pattern over this band, and will show a signal gain of 7 to 8 db. See figure 5 for design figures for 3-element arrays. Length of the Parasitic Elements www.americanradiohistory.com 492 THE Rotary Beams RADIO wavelength may be employed for greater front to -back ratios, but the radiation resistance of the array becomes quite low, the bandwidth of the array becomes very narrow, and the tuning becomes quite critical. Thus the Q of the antenna system will be increased as the spacing between the elements is decreased, and smaller optimum f r e q u e n c y coverage will result. z o Element Lengths When the parasitic element of a two -element array is used as a director, the following formulas may be used to determine the lengths of the driven element and the parasitic director, assuming an element diameter -to- length ratio of 200 to 400: 476 Driven element length (feet) - = = FAlc. Element spacing (feet) Figure = l 11c. 4 FIVE ELEMENT 28 MC BEAM ANTENNA AT W6SAI Antenna boom is made of twenty foot length of Sears, Roebuck Co. threeinch aluminum irrigation pipe. Spacing between elements is five feet. Elements are made of twelve foot lengths of 7/8 -inch aluminum tubing, with extension tips made of 3/4 -inch tubing. Gamma matching device, element clamps, and 'Oxen Yoke" element -toboom clamps are made by Continental Electronics 8 Sound Co., Dayton 27, Ohio. Beam dimensions are taken from figure 5. o.IS oz 0.2S ELEMENT SPACING (X) Figure 3 FRONT -TO -BACK RATIO AS A FUNCTION OF ELEMENT SPACING FOR A TWO -ELEMENT PARASITIC ARRAY Fmc. 450 Director length (feet) 0 I (PARASITIC ELEMENT TUNED FOR MAXIMUM GAIN) The effective bandwidth taken between the 1.5/1 standing wave points of an array cut to the above dimensions is about 2.5% of the operating frequency. This means that an array pre -cut to a frequency of 14,150 kilocycles would have a bandwidth of 350 kilocycles (plus or minus 175 kilocycles of the center frequency), and therefore would be effective over the whole 20 meter band. In like fashion, a 15 meter array should be pre -cut to 21,200 kilo- cycles. A beam designed for use on the 10 -meter band would have an effective bandwidth of some 700 kilocycles. Since the 10 -meter band is 1700 kilocycles in width, the array should either be cut to 28,500 kilocycles for operation in the low frequency portion of the band, or to 29,200 kilocycles for operation in the high frequency portion of the band. Operation of the antenna outside the effective bandwidth will increase the SWR on the transmission line, and noticeably degrade both the gain and front -to-back ratio performance. The height above ground also influences the F/B ratio. 25 -3 The Three -Element Array The three -element array using a director, driven element, and reflector will exhibit as much as 30 db front -to -back ratio and 20 db front -to -side ratio for low -angle radiation. The theoretical gain is about 9 db over a dipole in free space. In actual practice, the array will often show 7 to 10 db apparent gain over a horizontal dipole placed the same height above ground (at 28 and 14 Mc.). The use of more than three elements is desirable when the length of the supporting structure is such that spacings of approximately www.americanradiohistory.com Parasitic Arrays 491 SO 45 k 40 r 35 30 t .. . zs 20 ..... » Io s 0 1 0.15 0 2 ais ELEMENT SPACING (X) Figure 0 1 0.15 0 2 ELEMENT SPACING (X) 1 GAIN VS ELEMENT SPACING FOR A TWO ELEMENT CLOSE- SPACED PARASITIC BEAM ANTENNA WITH PARASITIC ELEMENT OPERATING AS A DIRECTOR OR REFLECTOR Such an antenna is capable of a signal gain of 5 db over a dipole, with a front -to -back ratio of 7 db to 15 db, depending upon the adjustment of the parasitic element. The parasitic element may be used either as a director or as a reflector. The optimum spacing for a reflector in a two -element array is approximately 0.13 wavelength and with optimum adjustment of the length of the reflector a gain of approximately 5 db will be obtained, with a feed -point resistance of about 25 ohms. If the parasitic element is to be used as a director the optimum spacing between it and the driven element is 0.11 wavelength. The gain will theoretically be slightly greater than with the optimum adjustment for a reflector (about 5.5 db) and the radiation resistance will be in the vicinity of 17 ohms. The general characteristics of a two -element parasitic array may be seen in figures 1, 2 and 3. The gain characteristics of a two -element array when the parasitic element is used as a director or as a reflector are shown. It can be seen that the director provides a maximum of 5.3 db gain at a spacing of slightly greater than 0.1 wavelength from the antenna. In the interests of greatest power gain and size conservation, therefore, the choice of a parasitic director would be wiser than the choice of a parasitic reflector, although the gain difference between the two is small. Figure 2 shows the relationship between the element spacing and the radiation resist- Figure 2 RADIATION RESISTANCE AS A FUNCTION OF ELEMENT SPACING FOR A TWO -ELEMENT PARASITIC ARRAY ance for the two element parasitic array for both the reflector and the director case. Since the optimum antenna-director spacing for maximum gain results in an antenna radiation resistance of about 17 ohms, and the optimum antenna- reflector spacing for maximum gain results in an antenna radiation resistance of about 25 ohms, it may be of advantage in some instances to choose the antenna with the higher radiation resistance, assuming other factors to be equal. Figure 3 shows the front -to -back ratio for the two element parasitic array for both the reflector and director cases. To produce these curves, the elements were tuned for maximum gain of the array. Better front -to -back ratios may be obtained at the expense of array gain, if desired, but the general shape of the curves remains the same. It can be readily observed that operation of the parasitic element as a reflector produces relatively poor front -toback ratios except when the element spacing is greater than 0.15 wavelength. However, at this element spacing, the gain of the array begins to suffer. Since a radiation resistance of 17 ohms is not unduly hard to match, it can be argued that the best all- around performance may be obtained from a two element parasitic beam employing 0.11 element spacing, with the parasitic element tuned to operate as a director. This antenna will provide a forward gain of 5.3 db, with a front -to -back ratio of 10 db, or slightly greater. Closer spacing than 0.11 www.americanradiohistory.com CHAPTER TWENTY -FIVE Rotary Beams The rotatable antenna array has become almost standard equipment for operation on the 28 -Mc. and 50 -Mc. bands and is commonly used on the 14-Mc. and 21 -Mc. bands and on those frequencies above 144 Mc. The rotatable array offers many advantages for both military and amateur use. The directivity of the antenna types commonly employed, particularly the unidirectional arrays, offers a worthwhile reduction in interference from undesired directions. Also, the increase in the ratio of low angle radiation plus the theoretical gain of such arrays results in a relatively large increase in both the transmitted signal and the signal intensity from a station being received. A significant advantage of a rotatable antenna array in the case of the normal station is that a relatively small amount of space is required for erection of the antenna system. In fact, one of the best types of installation uses a single telephone pole with the rotating structure holding the antenna mounted atop the pole. To obtain results in all azimuth directions from fixed arrays comparable to the gain and directivity of a single rotatable three- element parasitic beam would require several acres of surface. There are two normal configurations of radiating elements which, when horizontally polarized, will contribute to obtaining a low angle of radiation. These configurations are the end fire array and the broadside array. The con- ventional three- or four -element rotary beam may properly be called a unidirectional parasitic end-fire array, and is actually a type of yagi array. The flat -top beam is a type of bidirectional end-lire array. The broadside type of array is also quite effective in obtaining low -angle radiation, and although widely used in FM and TV broadcasting has seen little use by amateur stations in rotatable arrays. 25 -1 Unidirectional Parasitic End -Fire Arrays (Yogi Type) If a single parasitic element is placed on one side of a driven dipole at a distance of from 0.1 to 0.25 wavelength the parasitic element can be tuned to make the array substantially unidirectional. This simple array is termed parasitic beam. 25 -2 a two element The Two Element Beam The two element parasitic beam provides the greatest amount of gain per unit size of any array commonly used by radio amateurs. 490 www.americanradiohistory.com HANDBOOK Parasitic Arrays VHF 489 DRILL HOLES THROUGH BOOM AND PASS ELEMENTS THROUGH HOLES BOOM LENGTH DRIVEN ELEMENT GAIN= 16.1 ELEMENT DIMENSIONS REFLECTOR DIRECTORS 144 MC. 145 MC. 146 MC. 147 MC. 41^ 4ot 4°4' 404-- 36 367 ^ 24 . DIAM. If DB 2 METER BAND LENGTH ELEMENT (DIA M. I /6 -) = 36a SPACING FROM DIPOLE 19 D1= 7 02= 14.5 DRIVEN ELEMENT 36.5 D3= z2^ D4= 36 DS= 70. De= loz De= 134 lee. D9= 19e D7= e WIRE FOR 300 (1 MATCH. *10 WIRE MATCH FOR INSULATING PLATE 4500 BLATTEN TUBING AT ENDS. D10=230" D11=242" Figure DESIGN DIMENSIONS FOR A 2-METER "LONG YAGI" ANTENNA other hand, if a Yagi array of the same approximate size and weight as another antenna type is built, it will provide a higher order of power gain and directivity than that of the other antenna. The power gain of a Yagi antenna increases directly with the physical length of the array. The maximum practical length is entirely a mechanical problem of physically supporting the long series of director elements, although when the array exceeds a few wavelengths in length the element lengths, spacings, and Q's become more and more critical. The effectiveness of the array depends upon a proper combination of the mutual coupling loops between adjacent directors and between the first director and the driven element. On the Practically all work on Yagi antennas with more than three or four elements has been on an experimental, cut- and -try basis. Figure 19 19 provides dimensions for a typical Long Yagi antenna for the 2 -meter VHF band. Note that all directors have the same physical length. If the long Yagi is designed so that the directors gradually decrease in length as they progress from the dipole bandwidth will be increased, and both side lobes and forward gain will be reduced. One advantage gained from staggered director length is that the array can be shortened and lengthened by adding or taking away directors without the need for retuning the remaining group of parasitic elements. When all directors are the same length, they must be all shortened en masse as the array is lengthened, and viceversa when the array is shortened. A full discussion of Long Yagi antennas, including complete design and construction information may be had in the VHF Handbook, available through Radio Publications, Inc., Wilton., Conn. www.americanradiohistory.com 488 V -H -F and U -H -F THE Antenn as RADIO WOODBLOCK -BRASS TUBING A STUB C Figure 16 THE MOUNTING BLOCK FOR EACH SET OF ELEMENTS TRANSFORMER -B SHORTING BAR -C These tubes are welded onto the center tube of each group of three horizontal bracing tubes, and are so located to support the horizontal dipole at its exact center. The dipoles are attached to the supporting rods by means of small phenolic insulating blocks, as shown in figure 16. The radiators are therefore insulated from the screen reflector. The inner tips of the radiators are held by small polystyrene blocks for rigidity, and are cross connected to each other by a transposed length of TV -type 400 ohm open wire line. The entire array is fed at the point A -A, illustrated in figure 15. The matching system for the beam is mounted behind the reflector screen, and is shown in figure 17. A quarter -wave transformer (B) drops the relatively high impedance of the antenna array to a suitable value for the low impedance balun (D). An adjustable matching stub (C) and two variable capacitors (C, and C2) are employed for impedance matching. The two variable capacitors are mounted in a I C1&C2 =SO LUF WATERTIGHT COMPARTMENT APPROX r BALUN- D -- SNORTING BAR - D COPPER TUBING 72 R COAX CABLE Figure 17 THE MATCHING UNIT IN DETAIL FOR THE PE1PL BEAM DESIGN, WHICH ALLOWS THE USE OF 72 -OHM COAX watertight box, with the balun and matching stubs entering the bottom and top of the box, respectively. The matching procedure is carried out by the use of a standing wave meter (SWR bridge). A few watts of power are fed to the array through the SWR meter, and the setting of the shorting stub on C and the setting of the two variable capacitors are adjusted for lowest SüR at the chosen operating frequency. The capacity settings of the two variable capacitors should be equal. The final adjustment is to set the shorting stub of the balun (D) to remove any residual reactance that might appear on the transmission line. üith proper adjust- Figure 18 HORIZONTAL RADIATION PATTERN OF THE PE1PL ARRAY. THE FRONT -TOBACK RATIO IS ABOUT 28 db IN AMPLITUDE, AND THE FORWARD GAIN APPROXIMATELY 15 db. ment, the VSWR of the array may be held to less than 1.5 to 1 over a 2 megacycle range of the 2 -meter band. The horizontal radiation pattern of this array is shown in figure 18. a given power gain, the Yagi antenna can be built lighter, more compact, and with less wind resistance than any other type. Long Yogi Antennas www.americanradiohistory.com For HANDBOOK VHF Parasitic Arrays 487 The ends of the folded dipoles are made in following manner: Drive a length of dowel into the short connecting lengths of aluminum tubing. Then drill down the center of the dowel the with a clearance hole for the connecting screw. Then shape the ends of the connecting pieces to fit the sides of the element ends. After assembly the junctions may be dressed with a file and sandpaper until a smooth fit is ob- tained. The mast used for supporting the array is a 30 -foot spliced 2 by 2. A large discarded ball bearing is used as the radial load bearing and guy -wire termination. Enough of the upper-mast corners were removed with a draw-knife to permit sliding the ball bearing down about 9 feet from the top of the mast. The bearing then was encircled by an assembly of three pieces of dural ribbon to form a clamp, with ears for tightening screws and attachment of the guy wires. The bearing then was greased and covered with a piece of auto inner tube to serve as protection from the weather. Another junk box bearing was used at the bottom of the mast as a thrust bearing. The main boom s were made from 34-inch aluminum electrical conduit. Any size of small tubing will serve for making the elements. Note that the main boom is mounted at the balance center and not necessarily at the physical center. The pivot bolt in the offset head should be tightened sufficiently that there will be adequate friction to hold the array in position. Then an additional nut should be placed on the pivot bolt as a lock. In connecting the phasing sections between the T- junction and the centers of the folded dipoles, it is important that the center conductors of the phasing sections be connected to the same side of the driven elements of the antennas. In other words, when the antenna is oriented for horizontal polarization and the center of the coaxial phasing section goes to the left side of the top antenna, the center conductor of the other coaxial phasing section should go to the left side of the bottom antenn a. The "Screen Beam" for 2 Meters This highly effective ro- tary array for the 144 Mc. amateur band was designed by the staff of the Experimental Physics Laboratory, The Hague, Netherlands for use at the 2 meter experimental station PEIPL. The array consists of 10 half wave radiators fed in phase, and arranged in two stacked rows of five radiators. 0.2 wavelength behind this plane of radiators is a reflector screen, measuring approximately 15' x 9' in size. The antenna provides a power gain of 15 db, and a front to back ratio of approximately 28 db. ALL JOINTS WELDED Figure 15 DETAIL OF LAYOUT AND OF BEAM ASSEMBLY OF DIMENSIONS PEIPL The 10 dipoles are fed in phase by means of a length of balanced transmission line, a quarter -wave matching transformer, and a balun. A 72 -ohm coaxial line couples the array to the transmitter. A drawing of the array is shown in figure 15. The reflecting screen measures 14' 9" high by 8' 4" wide, and is made of welded %" diameter steel tubing. Three steel reinforcing bars are welded horizontally across the framework directly behind each pair of horizontal dipoles. The intervening spaces are filled with lengths of no. 12 enamel- coated copper wire to complete the screen. The spacing between the wires is 2 ". Four cross braces are welded to the corners of the frame for additional bracing, and a single vertical %" rod runs up the middle of the frame. The complete, welded frame is shown in figure 15. The no. 12 screening wires are run between 6-32 bolts placed in holes drilled in each outside vertical member of the frame. The antenna assembly is supported away from the reflector screen by means of ten lengths of % " steel tubing, each l' 3%4" long. www.americanradiohistory.com 486 V -H -F and U -H -F THE Antennas RADIO Figure 14 THE EIGHTELEMENT 144 -MC. ARRAY IN A HORIZONTAL POSITION appropriate cord. Hence, the operation is based on the offset head sketched in figure 13. Although a wood mast has been used, the same system may be used with a pipe mast. The 40 -inch lengths of RG -59/U cable (electrically 3i4 wavelength) running from the center of each folded dipole driven element to the coaxial T- junction allow enough slack to permit free movement of the main boom when changing polarity. Type RG -8 /U cable is run from the T- junction to the operating position. Measured standing -wave ratio was less than 2:1 over the 144 to 148 Mc. band, with the lengths and spacings given in figure 13. Construeion of the Array Most of the constructional aspects of the antenna array are self- evident from figure 13. However, the pointers given in the following paragraphs will be of assistance to those wishing to reproduce the array. The drilling of holes for the small elements should be done carefully on accurately marked centers. A small angular error in the drilling of these holes will result in a considerable misalignment of the elements after the array is assembled. The same consideration is true of the filing out of the rounded notches in the ends of the main boom for the fitting of the two antenna booms. Short lengths of wood dowel are used freely in the construction of the array. The ends of the small elements are plugged with an inch or so of dowel, and the ends of the antenna booms are similarly treated with larger discs pressed into place. www.americanradiohistory.com HANDBOOK VHF w- -,6 16 Parasitic Arrays 485 2ND DIRECTOR 35 5" I ___1ST DIRECTOR 36" RADIATOR 35" REFLECTOR 40" 5 RING BOLT ß_4B 60- TO FIT ELEMENTS -- O.D. - BOOM RG -59 /U EACH MAIN BOOMS- S- APPROX. BOOM FILE END CABLES 40 LONG -B /U CABLE ' T. COAXIAL FITTING RG TO SHAPE ENDS OF SHORT PIECES TO FIT CONTOUR INSULATING ROO. ENDS CUT DOWN TO GO INTO TUBING ABOU ENDS OF TUBING WOOD DOWELS INSIDE FOR STRENGTH TERMINALS - AS SHOWN. ANTENNA IS HORIZONTALLY POLARIZED PULL TO SWING MAIN BOOM FOR VERTICAL POLARITY. 90 CONTROL CORDS RG -13/U TO RIG ' 2X2 ROTATABLE CA WOOD MAST RADIAL BEARING CONSTRUCTIONAL DRAWING OF AN Figure 13 EIGHT- ELEMENT quency range. Although polarization has been loosely standardized in various areas of the country, exceptions are frequent enough so that it is desirable that the polarization of antenna radiation be easily changeable from horizontal to vertical. The antenna illustrated has shown a signal gain of about 11 db, representing a power gain of about 13. Although the signal gain of the TIPPABLE 144 -MC. ARRAY antenna is the same whether it is oriented for vertical or horizontal polarization, the horizontal beam width is smaller when the antenna is oriented for vertical polarization. Conversely, the vertical pattern is the sharper when the antenna system is oriented for horizontal polarization. The changeover from one polarization to the other is accomplished simply by pulling on the www.americanradiohistory.com 484 V -H -F and U -H sistors in series. If 2 -watt resistors are employed, this termination also is suitable for transmitter outputs of 10 watts or less. For higher powers, however, resistors having greater dissipation with negligible reactance in the upper v -h -f range are not readily available. For powers up to several hundred watts a suitable termination consists of a "lossy" line consisting of stainless steel wire (corresponding to no. 24 or 26 B &S gauge) spaced 2 inches, which in turn is terminated by two 390 -ohm 2 -watt carbon resistors. The dissipative line should be at least 6 wavelengths long. FOR 1 D=22^ ¡ METERS D-A-9 A-23} R-y R- REFLECTOR 40 LONG MWÚS2- BEND RADIUS 1B - FEED LINE TNRU HOLE, MIP-D 1X2 WELL-SEASONED WOOD GAIN 7.5 DB (20. LONG FOR 6/32 f 34 LONG FOR 11- METERS) SCREWS TAIL DIRECTOR 36" LONG Y 1) 75 TV LINE The rotary multi -element beam is undoubtedly the most popular type of v -h-f antenna in use. In general, the design, assembly and tuning of these antennas follows a pattern similar to the iarger types of rotary beam antennas used on the lower frequency amateur bands. The characteristics of these low frequency beam antennas are discussed in the next chapter of this Handbook, and the information contained in that chapter applies in general to the v -h -f beam antennas discussed herewith. simplest v -h -f beam for the beginner is the three -element Yagi array illustrated in figure 12. Dimensions are given for Yagis cut for the 2 -meter and ISmeter bands. The supporting boom for the Yagi may be made from a smoothed piece of 1" x 2" wood. The wood should be reasonably dry and should be painted to prevent warpage from exposure to sun and rain. The director and reflector are cut from lengths of %" copper tubing, obtainable from any appliance store that does service work on refrigerators. They should be cut to length as noted in figure 12. The elements should then be given a coat of aluminum paint. Two small holes are drilled at the center of the reflector and director and these elements are bolted to the wood boom by means of two wood screws. These screws should be of 111 the plated, or rust -proof variety. The driven element is made of a 78" length of ia" copper tubing, the ends bent back upon each other to form a folded dipole. If the tubing is packed with fine sand and the bending points heated over a torch, no trouble will be had in the bending process. If the tubing does collapse when it is bent, the break may be reThe Element Beam Antenna a .il WOOD BOOM Beam Antennas paired with _ L N3ULATING_BLbC5 Multi- Element V -H -F A Simple Three D A-DRIVEN ELEMENT 'LONG FLATTEN ENDS OF TUBING AND DRILL 24 -8 RADIO THE Antennas -F heavy -duty soldering iron. The Figure 12 SIMPLE 3- ELEMENT BEAM FOR 1'/ METERS 2 AND driven element is next attached to the center of the wood boom, mounted atop a small insulating plate made of bakelite, micarta or some other non -conducting material. It is held in place in the same manner as the parasitic elements. The two free ends of the folded dipole are hammered flat and drilled for a 6 -32 bolt. These bolts pass through both the insulating block and the boom, and hold the free tips of the element in place. A length of 75 -ohm Twin -Lead TV -type line should be used with this beam antenna. It is connected to each of the free ends of the folded dipole. If the.antenna is mounted in the vertical plane, the 75-ohm line should be brought away from the antenna for a distance of four to six feet before it drops down the tower to lessen interaction between the antenna elements and the feed line. The complete antenna is light enough to be turned by a TV rotator. A simple Yagi antenna of this type will provide a gain of 7 db over the entire 2 -meter or IS-meter band, and is highly recommended as an "easy -to- build" beam for the novice or beginner. An 8- Element "Tippoble" Array Figures 13 and 14 illustrate an 8- element rotary array for use on the 144 Mc. amateur band. This array is "tippable" to obtain either horizontal for 144 Mc. or vertical polarization. It is necessary that the transmitting and receiving station use the same polarization for the ground -wave signal propagation which is characteristic of this fre- www.americanradiohistory.com HANDBOOK VHF Rhombic 483 TOP VIEW 0' TILT ANGLE 4). 6A 131. Figure 10 RHOMBIC ANTENNA The optimum tilt NON -INDUCTIVE h SIDE LENGTH, S V -H -F RI, R22390 OHMS EACH 1OA Figure DESIGN CHART angle (see figure V -H -F 11 RHOMBIC ANTENNA CONSTRUCTION for zero-angle" radiation depends upon the length of the sides. 11) 10 to 16 db gain with a simpler construction than does a phased dipole array, and has the further advantage of being useful over a wide frequency range. Except at the upper end of the v -h -f range a rhombic array having a worthwhile gain is too large to be rotated. However, in locations 75 to 150 miles from a large metropolitan area a rhombic array is ideally suited for working into the city on extended (horizontally polarized) ground-wave while at the same time making an ideal antenna for TV reception. The useful frequency range of a v -h -f rhombic array is about 2 to I, or about plus 40% and minus 30% from the design frequency. This coverage is somewhat less than that of a high frequency rhombic used for sky -wave communication. For ground -wave transmission or reception the only effective vertical angle is that of the horizon, and a frequency range greater than 2 to I cannot be covered with a rhombic array without an excessive change in the vertical angle of maximum radiation or response. The dimensions of a v -h -f rhombic array are determined from the design frequency and figure 10, which shows the proper tilt angle (see figure 11) for a given leg length. The gain of a rhombic array increases with leg length. There is not much point in constructing a v -h -f rhombic array with legs shorter than about 4 wavelengths, and the beam width begins to become excessively sharp for leg lengths greater than about 8 wavelengths. A leg length of 6 wavelengths is a good compromise between beam width and gain. The tilt angle given in figure 10 is based upon a wave angle of zero degrees. For leg lengths of 4 wavelengths or longer, it will be necessary to elongate the array a few per cent (pulling in the sides slightly) if the horizon elevation exceeds about 3 degrees. Table I gives dimensions for two dual purpose rhombic arrays. One covers the 6-meter amateur band and the "low" television band. The other covers the 2 -meter amateur band, the "high" television band, and the 1%4-meter amateur band. The gain is approximately 12 db over a matched half wave dipole and the beam width is about 6 degrees. The Feed Line The recommended feed line is an open -wire line having a surge impedance between 450 and 600 ohms. With such a line the VSWR will be less than 2 to 1. A line with two -inch spacing is suitable for frequencies below 100 Mc., but one inch spacing (such as used in the Gonset Line for TV installations) is recommended for higher frequencies. The Termination If the array is to be used only for reception, a suitable termination consists of two 390 -ohm carbon re- 6 METERS AND LOW TV S 166' L (length) W -6 METERS 32' 10" 67' 4" (Width) METERS, NIGH BAND TV, AND 2 11Q 90. (side) S BAND 59' 4" 23' 11" warelenths at design frequency Tilt ongle 6B0 www.americanradiohistory.com TABLE I. 482 V -H -F and U -H -F THE Antennas RADIO A 450 -ONM TV LINE pA UHF HORN ANTENNA ANGLE BETWEEN SIDES OF MORN '"606 D OB 400 VHF HORIZONTALLY POLARIZED HORN Figure 8 TYPES OF HORN ANTENNAS The "two sided horn" of Figure BB may be fed by means of on open -wire transmission line. i ZA-A GAIN (DB) 430-OHM LINE 3 A 20 9 2a 390 1S TWO SIDES MADE OF WIRE MESH Figure 9 THE 60° HORN ANTENNA FOR USE ON FREQUENCIES ABOVE 144 MC. TWO Copper screen may also be used for the re- flecting planes. The values of spacing given in the corner reflector chart have been chosen such that the center impedance of the driven element would be approximately 70 ohms. This means that the element may be fed directly with 70 -ohm coaxial line, or a quarter-wave matching transformer such as a "Q" section may be used to provide an impedance match between the center- impedance of the element and a 460 -ohm line constructed of no. 12 wire spaced 2 inches. In many v -h -f antenna systems, waveguide transmission lines are terminated by pyramidal horn antennas. These horn antennas (figure 8A) will transmit and receive either horizontally or vertically polarized waves. The use of waveguides at 144 Mc. and 235 Mc., however, is out of the question because of the relatively large dimensions needed for a waveguide operating at these low frequencies. A modified type of horn antenna may still be used on these frequencies, since only one particular plane of polarization is of interest to the amateur. In this case, the horn antenna can be simplified to two triangular sides of the pyramidal horn. When these two sides are insulated from each other, direct excitation at the apex of the horn by a two-wire transmission line is possible. In a normal pyramidal horn, all four triangular sides are covered with conducting material, but when horizontal polarization alone is of interest (as in amateur work) only the vertical areas of the horn need be used. If vertical polarization is required, only the horizontal areas of the horn are employed. In either case, the system is unidirectional, away from the apex of the horn. A typical horn of this type is shown in figure 8B. The two metallic sides of the horn are insulated from each other, and the sides of the horn are made of small mesh "chicken wire" or copper window screening. A pyramidal horn is essentially a high -pass device whose low frequency cut-off is reached when a side of the horn is % wavelength. It will work up to infinitely high frequencies, the gain of the horn increasing by 6 db every time the operating frequency is doubled. The power gain of such a horn compared to a 1/2 wave dipole at frequencies higher than cutoff is: 8.4 A2 Power gain (db) A2 where A is the frontal area of the mouth of the horn. For the 60 degree horn shown in figure 8B the formula simplifies to: Power gain (db) = 8.4 D2, when pressed in terms of wavelength D is ex- When D is equal to one wavelength, the power gain of the horn is approximately 9 db. The gain and feed point impedance of the 60 degree horn are shown in figure 9. A 450 ohm open wire TV -type line may be used to feed the horn. 24 -7 VHF Horizontal Rhombic Antenna For v -h -f transmission and reception in a fixed direction, a horizontal rhombic permits www.americanradiohistory.com HANDBOOK Helical Antenna Beam 22 in. D .16%2 S in. 53 in. G Tubing o.d 1 in. The D and S dimensions are to the center of the tubing. These dimensions must be held rather closely, since the range from 144 through 225 Mc. represents just about the practical limit of coverage of this type of antenna system. DRIVEN DIPOLE SUPPORTING ME High -Band TV Coverage H Note that an array constructed with the above dimensions will give unusually good high -band TV reception in addition to covering the 144 Mc. and 220 -etc. amateur bands and the taxi and police services. On the 144 -Mc. band the beam width is approximately 60 degrees to the half -power points, while the power gain is approximately 11 db over a non -directional circularly polarized antenna. For high -band TV coverage the gain will be 12 to 14 db, with a beam width of about 50 degrees, and on the 220 -Mc. amateur band the beam width will be about 40 degrees with a power gain of approximately 15 db. The antenna system will receive vertically polarized or horizontally polarized signals with equal gain over its entire frequency range. Conversely, it will transmit signals over the same range, which then can be received with equal strength on either horizontally polarized vertically polarized receiving antennas. The standing -wave ratio will be very low over the complete frequency range if RG -63/U coaxial feed line is used. or 24 -6 481 The Corner -Reflector and Horn -Type Antennas The corner -reflector antenna is a good directional radiator for the v -h -f and u -h -f region. The antenna may be used with the radiating element vertical, in which case the directivity is in the horizontal or azimuth plane, or the system may be used with the driven element Figure 7 CONSTRUCTION OF THE "CORNER REFLECTOR" ANTENNA Such an antenna is capable of giving high gain with a minimum of complexity in the radiating system. It may be used either with horizontal or vertical polarization. Design data for the antenna is given in the Corner- Reflector Design Table. horizontal in which case the radiation is horizontally polarized and most of the directivity is in the vertical plane. With the antenna used as a horizontally polarized radiating system the array is a very good low -angle beam array although the nose of the horizontal pattern is still quite sharp. When the radiator is oriented vertically the corner reflector operates very satisfactorily as a direction -finding antenna. Design data for the corner -reflector antenna is given in figure 7 and in the chart Cosner Re /lector Design Data. The planes which make up the reflecting corner may be made of solid sheets of copper or aluminum for the u -h -f bands, although spaced wires with the ends soldered together at top and bottom may be used as the reflector on the lower frequencies. CORNER- REFLECTOR DESIGN DATA Corner Angle 90 60 60 60 60 Freq. Band, Mc. R 110" 110" 38" 24.5" 13" SO 50 144 220 420 NOTE: H S 82" 115" 140" 140" 40" 48" 25" 14" 30" 18" Refer to figure 7 A 200" 230" 100" 72" 36" Feed L G Imped. Approx. Gain, db 230" 230" 100" 72" 18" 18" 72 70 70 70 70 10 12 12 12 12 36" " 3" 5 for construction of corner- reflector an www.americanradiohistory.com 480 T and V -H -F U -H TRANSMIT RECEIVE ,,,,-ROUND OR SQUARE GROUND SCREEN L /\/\/\/\/\/\ G THE Antennas -F RADIO used at a single frequency or over a narrow band of frequencies, such as an amateur band. At the design frequency the beam width is about 50 degrees and the power gain about 12 db,referred to a non -directional circularly polarized antenna. For the frequency range 100 to 500 Mc. a suitable ground screen can be made from "chicken wire" poultry netting of -inch mesh, fastened to a round or square frame of either metal or wood. The netting should be of the type that is galvanized after weaving. A small, sheet metal ground plate of diameter equal to approximately D/2 should be centered on the screen and soldered to it. Tin, galvanized iron, or sheet copper. is suitable. The outer conductor of the RG -63/U (125 ohm) coax is connected to this plate, and the inner conductor contacts the helix through a hole in the center of the plate. The end of the coax should be taped with Scotch electrical tape to keep water out. The Ground Screen COAX FEED POINT (RG -63/U) AT CENTER OF GROUND SCREEN t D =+ 5= á G =oer. L. i. 1 A APPROX O.t1A CONDUCTOR DIA %= WAVELENGTH IN FREE SPACE Figure 6 THE "HELICAL BEAM" ANTENNA This type of directional antenna system gives excellent performance over o frequency range of 1.7 to 1.8 to 1. Its dimensions are such that it ordinarily is not practicable, however, for use as a rotatable array on frequencies below about 100 Mc. The center conductor of the feed line should pass through the ground screen for connection to the feed point. The outer conductor of the coaxial line should be grounded to the ground screen. the time of writing, there has been no standardization of the "twist" for general amateur work. Perhaps the simplest antenna configuration for a directional beam antenna having circular polarization is the helical beam popularized by Dr. John Kraus, W8JK. The antenna consists simply of a helix working against a ground plane and fed with coaxial line. In the u -h -f and the upper v -h -f range the physical dimensions are sufficiently small to permit construction of a rotatable structure without much difficulty. the dimensions are optimized, the characteristics of the helical beam antenna are such as to qualify it as a broad band antenna. An optimized helical beam shows little variation in the pattern of the main lobe and a fairly uniform feed point impedance averagWhen ing approximately 125 ohms over a frequency range of as much as 1.7 to 1. The direction of "electrical twist" (right or left handed) depends upon the direction in which the helix is wound. A six -turn helical beam is shown schematically in figure 6. The dimensions shown will give good performance over a frequency range of plus or minus 20 per cent of the design frequency. This means that the dimensions are not especially critical when the array is to be It should be noted that the beam proper consists of six full turns. The start of the helix is spaced a distance of S/2 from the ground screen, and the conductor goes directly from the center of the ground screen to the start of the helix. Aluminum tubing in the "SO" (soft) grade is suitable for the helix. Alternatively, lengths of the relatively soft aluminum electrical conduit may be used. In the v -h -f range it will be necessary to support the helix on either two or four wooden longerons in order to achieve sufficient strength. The longerons should be of as small cross section as will provide sufficient rigidity, and should be given several coats of varnish. The ground plane butts against the longerons and the whole assembly is supported from the balance point if it is to be rotated. Aluminum tubing in the larger diameters ordinarily is not readily available in lengths greater than 12 feet. In this case several lengths can be spliced by means of short telescoping sections and sheet metal screws. The tubing is close wound on a drum and then spaced to give the specified pitch. Note that the length of one comp e t e turn when spaced is somewhat greater than the circumference of a circle having the diameter D. The Helix 1 Broad -Band 144 to 225 Mc. Helical Beam A highly useful v -h -f helical beam which will receive sig- nals with good gain over the complete frequency range from 144 through 225 Mc. may be constructed by using the following dimensions (180 Mc. design center): www.americanradiohistory.com HANDBOOK Discone 0.1 479 Antenna D 400i 300 200 160 160 140 120' 110 MO' 90 so 50n. COAX (PIG-4/U, ETC.) To 60 Figure SA THE ''DISCONE " BROAD -BAND RADIATOR This antenna system radiates a vertically polarized wave over a very wide frequency range. The "disc" may be made of solid met al sheet, a group of radials, or wire screen; the "cone" may best be constructed by forming a sheet of thin aluminum. A single antenna may be used for operation on the 50, 144, and 220 Mc. amdteur bands. The dimension D is determined by the lowest frequency to be employed, and is given in the 50 O.! t 0 15 DIN DESIGN 2 2.5 3 4 6 FEET Figure 5B CHART FOR THE "DISCONE" ANT ENN A of the skirt directly to an effective ground plane such as the top of an automobile. chart of figure 58. 24 -5 Helical Beam Antennas VSXRof less than 1.5 will be obtained throughout the operating range of the antenna. The Discone antenna may be considered as a cross between an electromagnetic horn and an inverted ground plane unipole antenna. It looks to the feed line like a properly terminated high -pass filter. Construction Details The top disk and the conical skirt may be fabricated either from sheet metal, screen (such as "hardware cloth "), or 12 or more "spine" radials. If screen is used a supporting framework of rod or tubing will be necessary for mechanical strength except at the higher frequencies.. If spines are used, they should be terminated on a ring for mechanical strength except at the higher frequencies. The top disk is supported by means of three insulating pillars fastened to the skirt. Either polystyrene or low -loss ceramic is suitable for the purpose. The apex of the conical skirt is grounded to the supporting mast and to the outer conductor of the coaxial line. The line is run down through the supporting mast. An alternative arrangement, one suitable for certain mobile applications, is to fasten the base stiff Most v -h -f and u -h -f antennas are either verpolarized or horizontally polarized (plane polarization). However, circularly polarized antennas have interesting characteristics which may be useful for certain applications. The installation of such an antenna can effectivèly solve the problem of horizontal vs. vertical polarization. A circularly polarized wave has its energy divided equally between a vertically polarized component and a horizontally polarized component, the two being 90 degrees out of phase. The circularly polarized wave may be either "left handed" or "right handed," depending upon whether the vertically polarized component leads or lags the horizontal component. A circularly polarized antenna will respond to any plane polarized wave whether horizontally polarized, vertically polarized, or diagonally polarized. Also, a circular polarized wave can be received on a plane polarized antenna, regardless of the polarization of the latter. When using circularly polarized antennas at both ends of the circuit, however, both must be left handed or both must be right handed. This offers some interesting possibilities with regard to reduction of QRM. At tically www.americanradiohistory.com 478 V -H -F and U -H -F Antennas THE f 1 36" TYP. RADIO jALUMINUM TUBING . 2X2 191 IR' TYP. 220 MC. w 19" T I S.I. = ak- E/r, ..UU/MUMCM7SSBAB TIGHTENS IT UP. 300-OHM FEEDLIN TOP APEX CONNECTS TO INNER CONNECTOR LOWER APEX CONNECTS TO OUTER CONDUCTOR APICES FORMED ~ -OF SHEET METAL 300-OHM TUBULAR TWIN LEAD 20' 300-OHM FEEDLINE RD-B /U CABLE Figure 3 THE DOUBLE SKELETON CONE ANTENNA A skeleton cone has been substituted for the single element radiator of figure 2C. This greatly increases the bandwidth. If at least 10 elements are used for each skeleton cone and the angle of revolution and element length are optimized, a low SWR con be obtained over o frequency range of at least two octaves. To obtain this order of bandwidth, the element length L should be approximately 0.2 wavelength at the lower frequency end of the band, and the angle of revolution optimized for the lowest maximum VSWR within the frequency range to be covered. A greater improvement in the impedance -frequency characteristic can be achieved by adding elements than by increasing the diameter of the elements. With only 3 elements per "cone'. and a much smaller angle of revolution a low SWR can be obtained over a frequency range of approximately 1.3 to 1.0 when the element lengths are optimized. Figure 4 NONDIRECTIONAL ARRAYS FOR use. acing frequency. The antenna then will perform well over a frequency range of at least At certain frequencies within this range the vertical pattern will tend to "lift" slightly, causing a slight reduction in gain at zero angular elevation, but the reduction is very slight. Below the frequency at which the slant height of the conical skirt is equal to a free space quarter wavelength the standing-wave ratio starts to climb, and below a frequency approximately 20 per cent lower than this the standing -wave ratio climbs very rapidly. This is termed the cut off frequency of the antenna. By making the slant height approximately equal to a free -space quarter wavelength at the lowest frequency employed (refer to chart), a 8 to 1. work over several octaves, the gain varying only slightly over a very wide frequency range. Commercial versions of the Discone antenna for various applications are manufactured by the Federal Telephone and Radio Corporation. A Discone type antenna for amateur work can be fabricated from inexpensive materials with ordinary hand tools. A Discone antenna suitable for multi -band amateur work in the v- h /u-h -f range is shown schematically in figure 5A. The distance D should be made approximately equal to a free space quarter wavelength at the lowest oper- 144 MC. AND 235 MC. On right is shown two band installation. The whole system may easily be dissembled and carried on a ski -rock atop a car for portable www.americanradiohistory.com Vertically Polarized Arrays HANDBOOK CLOSED I r ~OPEN Figure 2 THREE VERTICALLY -POLARIZED LOW -ANGLE RADIATORS Shown at (A) is the "sleeve" or "'hypodermic" type of radiator. At (©) is shown the ground-plane vertical, and (C) shows a modification of this antenna system which increases the feed-point impedance to a value such that the system may be fed directly from o coaxial line with no standing waves on the feed line. matching transformer, and a good match is obtained. In actual practice the antenna would consist of a quarter-wave rod, mounted by means of insulators atop a pole or pipe mast. Elaborate insulation is not required, as the voltage at the lower end of the quarter -wave radiator is very low. Self- supporting rods from 0.25 to 0.28 wavelength would be extended out, as in the illustration, and connected together. As the point of connection is effectively at ground potential, no insulation is required; the horizontal rods may be bolted directly to the supporting pole or mast, even if of metal. The coaxial line should be of the low loss type especially designed for v -h -f use. The outside connects to the junction of the radials, and the inside to the bottom end of the vertical radiator. An antenna of this type is moderately simple to construct and will give a good account of itself when fed at the lower end of the radiator directly by the 52 -ohm RG -8 /U coaxial cable. Theoretically the standing -wave ratio will be approximately 1.5-to -1 but in practice this moderate s -w -r produces no deleterious effects, even on coaxial cable. The modification shown in figure 2C permits matching to a standard 50- or 70 -ohm flexible coaxial cable without a linear transformer. If the lower rods hug the line and supporting mast 477 rather closely, the feed -point impedance is about 70 ohms. If they are bent out to form an angle of about 30° with the support pipe the impedance is about 50 ohms. The number of radial legs used in a ground plane antenna of either type has an important effect on the feed -point impedance and upon the radiation characteristics of the antenna system. Experiment has shown that three radials is the minimum number that should be used, and that increasing the number of radials above six adds substantially nothing to the effectiveness of the antenna and has no effect on the feed -point impedance. Experiment has shown, however, that the radials should be slightly longer than one -quarter wave for best results. A length of 0.28 wavelength has been shown to be the optimum value. This means that the radials for a 50 -Mc. ground -plane vertical antenna should be 65" in length. The bandwidth of the antenna of figure 2C can be increased considerably by substituting several space -tapered rods for the single radiating element, so that the "radiator" and skirt are similar. If a sufficient number of rods are used in the skeleton cones and the angle of revolution is optimized for the particular type of feed line used, this antenna exhibits a very low SWR over a 2 to 1 frequency range. Such an arrangement is illustrated schematically in figure 3. Double Skeleton Cone Antenna Nondirectional Vertical Array Half-wave elements may be stacked in the vertical plane to provide a non -directional pattern with good horizontal gain. An array made up of four half -wave vertical elements is shown in figure 4A. This antenna provides a circular pattern with a gain of about 4.5 db over a vertical dipole. It may be fed with 300 -ohm TV -type line. The feedline should be conducted in such a way that the vertical portion of the line is at least one-half wavelength away from the vertical antenna elements. A suitable mechanical assembly is shown in figure 4B for the 144 -Mc. and 235 -Mc. amateur bands. A 24 -4 The Discone Antenna The Discone antenna is a vertically polarized omnidirectional radiator which has very broad band characteristics and permits a simple, rugged structure. This antenna presents a substantially uniform feed -point impedance, suitable for direct connection of a coaxial line, over a range of several octaves. Alsg, the vertical pattern is suitable for ground -wave www.americanradiohistory.com 476 V -H -F and U -H -F RADIO THE Antennas 1 1 2=70n VECTOR SUM OF 2 PATTERNS COAXIAL LINE TO TRANSMITTER LOW Z TRANSMISSION LINE TO XMTR O Figure HORIZONTALLY © 1 THREE NONDIRECTIONAL, radiation at the very low elevation angles are not recommended for v -h -f and u -h -f work. It is for this reason that the horizontal dipole and horizontally- disposed colinear arrays are generally unsuitable for work on these frequencies. Arrays using broadside or end-fire elements do concentrate radiation at low elevation angles and are recommended for v -h-f work. Arrays such as the lazy -H, Sterba curtain, flat -top beam, and arrays with parasitically excited elements are recommended for this work. Dimensions for the first three types of arrays may be determined from the data given in the previous chapter, and reference may be made to the Table of Wavelengths given in this chapter. Arrays using vertically- stacked horizontal dipoles, such as are used by commercial television and FM stations, are capable of giving high gain without a sharp horizontal radiation pattern. If sets of crossed dipoles, as shown in figure 1A, are fed 90° out of phase the resulting system is called a turnstile antenna. The 90° phase difference between sets of dipoles may be obtained by feeding one set of dipoles with a feed line which is one -quarter wave longer than the feed line to the other set of dipoles. The field strength broadside to one of the dipoles is equal to the field from that dipole alone. The field strength at a point at any other angle is equal to the vector sum of the fields from the two dipoles at that angle. A nearly circular horizontal pattern is produced by this antenna. A second antenna producing a uniform, horizontally polarized pattern is shown in figure 1B. This antenna employs three dipoles bent to form a circle. All dipoles are excited in phase, and are center fed. A bazooka is included in the system to prevent unbalance in the coaxial feed system. POLARIZED ANTENNAS A third nondirectional antenna is shown in figure IC. This simple antenna is made of two half-wave elements, of which the end quarter wavelength of each is bent back 90 degrees. The pattern from this antenna is very much like that of the turnstile antenna. The field from the two quarter -wave sections that are bent back are additive because they are 180 degrees out of phase and are a half wavelength apart. The advantage of this antenna is the simplicity of its feed system and construction. 24 -3 Simple Vertical -Polarized Antennas For general coverage with a single antenna, single vertical radiator is commonly employed. A two -wire open transmission line is not suitable for use with this type antenna, and coaxial polyethylene feed line such as RG-8 /U is to be recommended. Three practical methods of feeding the radiator with concentric line, with a minimum of current induced in the outside of the line, are shown in figure 2. Antenna (A) is known as the sleeve antenna, the lower half of the radiator being a large piece of pipe up through which the concentric feed line is run. At (B) is shown the ground plane vertical, and at (C) a modification of this latter antenna. resistance of the ground The radiation plane vertical is approximately 30 ohms, which is not a standard impedance for coaxial line. To obtain a good match, the first quarter wavelength of feeder may be of 52 ohms surge impedance, and the remainder of the line of approximately 75 ohms impedance. Thus, the first quarter -wave section of line is used as a a www.americanradiohistory.com HANDBOOK Antenna Polarization There is no point in using copper tubing for an antenna on the medium frequencies. The reason is that considerable tubing would be required, and the cross section still would not be a sufficiently large fraction of a wavelength to improve the antenna bandwidth characteristics. At very high and ultra high frequencies, however, the radiator length is so short that the expense of large diameter conductor is relatively small, even though copper pipe of 1 inch cross section is used. With such conductors, the antenna will tune much more broadly, and often a broad resonance characteristic is desirable. This is particularly true when an antenna or array is to be used over an entire amateur band. It should be kept in mind that with such large cross section radiators, the resonant length of the radiator will be somewhat shorter, being only slightly greater than 0.90 of a half wavelength for a dipole when heavy copper pipe is used above 100 Mc. Radiator Cross Section The matter of insulation is of prime importance at very high frequencies. Many insulators that have very low losses as high as 30 Mc. show up rather poorly at frequencies above 100 Mc. Even the low loss ceramics are none too good where the r -f voltage is high. One of the best and most practical insulators for use at this frequency is polystyrene. It has one disadvantage, however, in that it is subject to fracture and to deformation in the presence of heat. It is common practice to design v -h -f and u -h -f antenna systems so that the various radiators are supported only at points of relatively low voltage; the best insulation, obviously, is air. The voltages on properly operated untuned feed lines are not high, and the question of insulation is not quite so important, though insulation still should be of good grade. Insulation 475 TABLE OF WAVELENGTHS Fra. quency in Mc. t/4 Wove Free Space 50.0 50.5 51.0 51.5 52.0 52.5 53.0 54.0 58.5 57.9 57.4 56.8 56.3 55.7 54.7 144 145 146 147 148 235 236 237 238 239 240 Wave Anrenna 55.5 55.0 54.4 53.9 53.4 1/2 Wave Free Space 1/2 Wane An- renna 51 .4 109.5 111.0 109.9 108.8 107.8 106.7 105.7 104.7 102.8 20.5 20.4 20.2 20.0 19.9 19.2 18.9 18.8 18.6 41.0 40.8 40.4 40.0 39.9 38.5 38.3 38.0 37.6 37.2 12.6 12.5 12.5 12.4 12.4 12.3 11.8 11.8 11.7 11.7 11.6 11.6 25.2 25.1 25.0 24.9 24.8 24.6 23.6 23.5 23.5 23.4 23.3 23.2 14.1 13.25 S9.1 420 425 430 1/4 7.05 6.95 6.88 5 2. 8 52 4 19.1 6.63 6.55 6.48 118.1 116.9 115.9 114.7 113.5 112.5 1 1 1 .5 13.9 13.8 13.1 12.95 All dimensions ore in inches. Lengths hove in most cases been rounded off to three significant figures. "1/2 -Wave Free -Space' column shown above should be used with Lecher wires for frequency measurement. . Antenna Commercial broadcasting in the U.S.A. for both FM and television in the v -h -f range has been standarized on horizontal polarization. One of the main reasons for this standardization is the fact that ignition interference is reduced through the use of a horizontally polarized receiving antenna. Amateur practice, however, is divided between horizontal and vertical polarization in the v -h-f and u -h -f range. Mobile stations are invariably vertical cally polarized due to the physical limitations imposed by the automobile antenna installation. Most of the stations doing intermittent or occasional work on these frequencies use a simple ground-plane vertical antenna for both transmission and reception. However, those Polarization stations doing serious work and striving for maximum -range contacts on the 50 -Mc. and 144 -Mc. bands almost invariably use horizon- tal polarization. Experience has shown that there is a great attenuation in signal strength when using crossed polarization (transmitting antenna with one polarization and receiving antenna with the other) for all normal ground -wave contacts on these bands. When contacts are being made through sporadic -E reflection, however, the use of crossed polarization seems to make no discernible difference in signal strength. So the operator of a station doing v -h -f work (particularly on the 50 -Mc. band) is faced with a problem: If contacts are to be made with all stations doing work on the same band, provision must be made for operation on both horizontal and vertical polarization. This problem has been solved in many cases through the construction of an antenna array that may be revolved in the plane of polarization in addition to being capable of .rotation in the azimuth plane. An alternate solution to the problem which involves less mechanical construction is simply to install a good ground -plane vertical antenna for all vertically- polarized work, and then to use a multi -element horizontally- polarized array for dx work. 24 -2 Simple Horizontally - Polarized Antennas Antenna systems which do not concentrate www.americanradiohistory.com 474 V -H -F and U -X -F THE Antennas RADIO that both are directed at the station being received. Many instances have been reported where a frequency band sounded completely dead with a simple dipole receiving antenna but when the receiver was switched to a three element or larger array a considerable amount of activity from 80 to 160 miles distant was heard. of about 50 feet or less. For longer runs, either the u -h -f or v -h -f TV open -wire lines may be used with good overall efficiency. The v -h -f line is satisfactory for use on the amateur 420 -Mc. band. Angle of connection, however, is the antenna changeover relay. Reflections at the antenna changeover relay become of increasing importance as the frequency of transmission is increased. When coaxial cable is used as the antenna transmission line, satisfactory coaxial antenna changeover relays with low reflection can be used. One type manufactured by Advance Electric & Relay Co., Los Angeles 26, Calif., will give a satisfactorily low value of reflection. On the 235-Mc. and 420 -Mc. amateur bands, the size of the antenna array becomes quite small, and it is practical to mount two identical antennas side by side. One of these antennas is used for the transmitter, and the other antenna for the receiver. Separate transmission lines are used, and the antenna relay may be eliminated. The useful portion of the signal in the v -h -f and u -h -f range for short or medium distance communication is that which is radiated at a very low angle with respect to the surface of the earth; essentially it is that signal which is radiated parallel to the surface of the earth. A vertical antenna transmits a portion of its radiation at a very low angle and is effective for this reason; its radiation is not necessarily effective simply because it is vertically polarized. A simple horizontal dipole radiates very little low-angle energy and hence is not a satisfactory v -h -f or u -h -f radiator. Directive arrays which concentrate a major portion of the radiated signal at a low radiation angle will prove to be effective radiators whether their signal is horizontally or vertically polarized. In all cases, the radiating system for v -h -f and u -h -f work should be as high and in the clear as possible. Increasing the height of the antenna system will produce a very marked improvement in the number and strength of the signals heard, regardless of the actual type of antenna used. Radiation Transmission lines to v -h -f and u -h -f antenna systems may be either of the parallel- conductor or coaxial conductor type. Coaxial line is recommended for short runs and closely spaced open -wire line for longer runs. Wave guides may be used under certain conditions for frequencies greater than perhaps 1500 Mc. but their dimensions become excessively great for frequencies much below this value. Non- resonant transmission lines will be found to be considerably more efficient on these frequencies than those of the resonant type. It is wise to to use the very minimum length of transmission line possible since transmission line losses at frequencies above about 100 Mc. mount very rapidly. Open sines should preferably be spaced closer than is common for longer wavelengths, as 6 inches is an appreciable fraction of a wavelength at 2 meters. Radiation from the line will be greatly reduced if 1 -inch or 11/4inch spacing is used, rather than the more common 6-inch spacing. Ordinary TV-type 300 -ohm ribbon may be used on the 2 -meter band for feeder lengths Transmission Lines It is recommended that the same antenna be used for transmitting and receiving in the v -h -f and u -h -f range. An ever- present problem in this Antenna Changeover vertical radiator for general coverage u -h -f use should be made either 1/4 or % wavelength long. Longer vertical antennas do not have their maximum radiation at right angles to the line of the radiator (unless co- phased), and, therefore, are not practicable for use where greatest possible radiation parallel to the earth is desired. Unfortunately, a feed system which is not perfectly balanced and does some radiating, not only robs the antenna itself of that much power, but distorts the radiation pattern of the antenna. As a result, the pattern of a vertical radiator may be so altered that the radiation is bent upwards slightly, and the amount of power leaving the an t e n n a parallel to the earth is greatly reduced. A vertical half -wave radiator fed at the bottom by a quarter -wave stub is a good example of this; the slight radiation from the matching section decreases the power radiated parallel to the earth by nearly 10 db. The only cure is a feed system which does not disturb the radiation pattern of the antenna itself. This means that if a 2 -wire line is used, the current and voltages must be exactly the same (though 180° out of phase) at any point on the feed line. It means that if a concentric feed line is used, there should be no current flowing on the outside of the outer conductor. Effect of Feed System on Radiation Angle www.americanradiohistory.com A CHAPTER TWENTY -FOUR V-li-F and U-li-F The very- high -frequency or v -h -f frequency range is defined as that range falling between 30 and 300 Mc. The ultra- high -frequency or u -h -f range is defined as falling between 300 and 3000 Mc. This chapter will be devoted to the design and construction of antenna systems for operation on the amateur 50 -Mc., 144 Mc., 235 -Mc., and 420 -Mc. bands. Although the basic principles of antenna operation are the same for all frequencies, the shorter physical length of a wave in this frequency range and the differing modes of signal propagation make it possible and expedient to use antenna systems different in design from those used on the range from 3 to 30 Mc. 24 -1 Antennas station. Even a much simpler and smaller three or four -element parasitic array having a gain of 7 to 10 db will produce a marked improvement in the received signal at the other station. 11 o w e v e r, as all v -h -f and u -h -f workers know, the most important contribution of a high -gain antenna array is in reception. If a remote station cannot be heard it obviously is impossible to make contact. The limiting factor in v -h -f and u -h -f reception is in almost every case the noise generated within the receiver itself. Atmospheric noise is almost nonexistent and ignition interference can almost invariably be reduced to a satisfactory level through the use of an effective noise limiter. Even with a grounded -grid or neutralized triode first stage in the receiver the noise contribution of the first tuned circuit in the receiver will be relatively large. Hence it is desirable to use an antenna system which will deliver the greatest signal voltage to the first tuned circuit for a given field strength at the receiving location. Antenna Requirements Any type of antenna system useable on the lower frequencies may be used in the v -h -f and u -h -f bands. In fact, simple non -directive half wave or quarter -wave vertical antennas are very popular for general transmission and reception from all directions, especially for short-range work. But for serious v -h -f or u -h -f work the use of some sort of directional antenna array is a necessity. In the first place, when the transmitter power is concentrated into a narrow beam the apparent transmitter power at the receiving station is increased many times. A "billboard" array or a Sterba curtain having a gain of 16 db will make a 25 -watt transmitter sound like a kilowatt at the other Since the field intensity being produced at the receiving location by a remote transmitting station may be assumed to be constant, the receiving antenna which intercepts the greatest amount of wave front, assuming that the polarization and directivity of the receiving antenna is proper, will be the antenna which gives the best received signal -to-noise ratio. An antenna which has two square wavelengths effective area will pick up twice as much signal power as one which has one square wavelength area, assuming the same general type of antenna and 473 www.americanradiohistory.com 472 High Frequency Directive Antennas Thus it is seen that, while maximum gain occurs with two stacked dipoles at a spacing of about 0.7 wavelength and the space directivity gain is approximately 5 db over one element under these conditions; the case of two flat top or parasitic arrays stacked one above the other is another story. Maximum gain will occur at a greater spacing, and the gain over one array will not appreciably exceed 3 db. When two broadside curtains are placed one ahead of the other in end -fire relationship, the aggregate mutual impedance between the two curtains is such that considerable spacing is required in order to realize a gain approaching 3 db (the required spacing being a function of the size of the curtains). While it is true that a space directivity gain of approximately 4 db can be obtained by placing one, half -wave dipole an eighth wavelength ahead of another and feeding them 180 degrees out of phase, a gain of less than 1 db is obtained when the same procedure is applied to two large broadside curtains. To obtain a gain of approximately 3 db and retain a bidirectional pattern, a spacing of many wavelengths is required between two large curtains placed one ahead of the other. A different situation exists, however, when one driven curtain is placed ahead of an identical one and the two are phased so as to give a unidirectional pattern. When a unidirectional pattern is obtained, the gain over one curtain will be approximately 3 db regardless of the spacing. For instance, two large curtains placed one a quarter wavelength ahead of the other may have a space directivity gain of only 0.5 db over one curtain when the two are driven 180 degrees out of phase to give a bidirectional pattern (the type of pattern obtained with a single curtain). However, if they are driven in phase quadrature (and with equal currents) the gain is approximately 3 db. The directivity gain of a composite array also can be explained upon the basis of the directivity patterns of the component arrays alone, but it entails a rather complicated picture. It is sufficient for the purpose of this discussion to generalize and simplify by saying that the greater the directivity of an end fire array, the farther an identical array must be spaced from it in broadside relationship to obtain optimum performance; and the greater the directivity of a broadside array, the farther an identical array must be spaced from it in end -fire relationship to obtain optimum performance and retain the bidirectional charac- teristic. It is important to note that while a bidirectional end -fire pattern is obtained with two driven dipoles when spaced anything under a half wavelength, and while the proper phase relationship is 180 degrees regardless of the spacing for all spacings not exceeding one half wavelength, the situation is different in the case of two curtains placed in end -fire relationship to give a bidirectional pattern. For maximum gain at zero wave angle, the curtains should be spaced an odd multiple of one half wavelength and driven so as to be 180 degrees out of phase, or spaced an even multiple of one half wavelength and driven in the same phase. The optimum spacing and phase relationship will depend upon the directivity pattern of the individual curtains used alone, and as previously noted the optimum spacing increases with the size and directivity of the component arrays. A concrete example of a combination broadside and end -fire array is two Lazy H arrays spaced along the direction of maximum radiation by a distance of four wavelengths and fed in phase. The space directivity gain of such an arrangement is slightly less than 9 db. However, approximately the same gain can be obtained by juxtaposing the two arrays side by side or one over the other in the same plane, so that the two combine to produce, in effect, one broadside curtain of twice the area. It is obvious that in most cases it will be more expedient to increase the area of a broadside array than to resort to a combination of end fire and broadside directivity. One exception, of course, is where two curtains are fed in phase quadrature to obtain a unidirectional pattern and space directivity gain of approximately 3 db with a spacing between curtains as small as one quarter wavelength. Another exception is where very low angle radiation is desired and the maximum pole height is strictly limited. The two aforementioned Lazy H arrays when placed in end-fire relationship will have a considerably lower radiation angle than when placed side by side if the array elevation is low, and therefore may under some conditions exhibit appreciably more practical signal gain. www.americanradiohistory.com HANDBOOK Triplex Beam 471 ' RO.[ " Figure 24 THE TRIPLEX FLATTOP BEAM ANTENNA FOR 10, 15 AND 20 METERS S u- 11 MAXIMUM MAX. RADIATION RADIATION 4.S DS 4.5 3000 LINE OR TO TRANSMITTER ANY LENGTH DIMENSIONS to one -quarter wave spacing may be used on the fundamental for the one -section types and also the two -section center-fed, but it is not desirable to use more than 0.15 wavelength spacing for the other types. Although the center -fed type of flat -top generally is to be preferred because of its symmetry, the end -fed type often is convenient or desirable. For example, when a flat -top beam is used vertically, feeding from the lower end is in most cases more convenient. If a multisection flat -top array is end -fed instead of center-fed, and tuned feeders are used, stations off the ends of the array can be worked by tying the feeders together and working the whole affair, feeders and all, as a long wire harmonic antenna. A single -pole double throw switch can be used for changing the feeders and directivity. The Triplex The Triplex beam is a modified of the W8] K antenna which uses folded dipoles for the half wave elements of the array. The use of folded dipoles results in higher radiation resistance of the array, and a high overall system performance. Three wire dipoles are used for the elements, and 300 -ohm Twin -Lead is Beam version 10M. 15M. 20M MATERIAL L 1'S 21'5' 32'2 iEL[tA[b 3' S 5'0. 7'11. II' D 7'2' 10'7' 14'4" 3000M RIeeON used for the two phasing sections. A recommended assembly for Triplex beams for 28 Mc., 21 Mc., and 14 Mc. is shown in figure 24. The gain of a Triplex beam is about 4.5 db over a dipole. 23 -8 Combination End -Fire and Broadside Arrays Any of the end -fire arrays previously described may be stacked one above the other or placed end to end (side by side) to give greater directivity gain while maintaining a bidirectional characteristic. However, it must be kept in mind that to realize a worthwhile increase in directivity and gain while maintaining a bidirectional pattern the individual arrays must be spaced sufficiently to reduce the mutual impedances to a negligible value. When two flat top beams, for instance, are placed one above the other or end to end, a center spacing on the order of one wavelength is required in order to achieve a worthwhile increase in gain, or approximately 3 db. www.americanradiohistory.com THE High Frequency Directive Antennas 470 CENTER FED RADIO END FED TO CENTER FLAT TOP or I.-A-4 It r MATCHING STUB M L, {-oi 1- SECTION 1- SECTION , L, 11 A --{ 1 -L2 -'"i 1-IX 2- SECTION -L2 g .---L3 01- L2-- t fol La 12M CONNCCF ATCr L3I l'L3 STUB ERS 3-SECTION r S M L2--1 Ioi I---La loi 4-SECTION g r S L3- L4 2M S M , I--- L3- 1-o- -- L3- 4pr.- L3-404-- L3- k-- L3-4 1_La _J-4- La ----1-61--1-3-.4 FIGURE 23 FLAT -TOP BEAM (8JK ARRAY) DESIGN DATA. FREQUENCY Spar. 'os 7.0-7.2 Mc. X/8 7.2 -7.3 14.0 -14.4 14.0 -14.4 14.0 -14.4 14.0 -14.4 28.0-29.0 28.0 -29.0 29.0 -30.0 29.0 -30.0 a/8 )/B A S L. L, 17'4' 34 17'0' 33'6' 8'8' 17' 10'5' 17' 13'11' 17' a/4 17'4' 17' .15) 5'2' 8'6' .15X .20X a/4 .15X X/4 8'8' 5'0' 8'4' 8'6' 8'3' 8'3' 60' 59' 30' 30' 30' 30' L, L. 52'8' 51'8' 44' 26'4 22' 43'1' M D 8'10' 4' 8'8' 4' 4'S' 2' 2S'3' 20' 5'4' 2' 22'10' 7'2' 2' 20'8' 8'10' 2' 15' 12'7' 10' 2'8' 1'6' 15' 10'4' 4'S' 1'6' 14'6' 12'2' 9'8' 2'7' 1'6' 14'6' 10'0' 4'4' 1'6' (/) A ('A) A (1/4) X 60' 59' 30' 29' 27' 25' 96' 94' 48' 47' 45' 43' 24' 22' 23' 21' 4' 4' approx. approx. approx. approx. 26' 26' 13' 12' 10' 8' 7' 5' 7' 5' 15' 13' 15' 13' 2' 2' 3' 4' 1' 2' 1' 2' Dimension chart for flat -top beam antennas. The meanings of the symbols are as fo lows: L. L. and L,, the lengths of the sides of the flat-top sections as shown. L, is length of the sides of single -section center -fed, L. single- section end -fed and 2- section center -fed, L, 4- section center -fed and end -sections of 4- section end -fed, and L, middle sections of 4- section end -fed. S, the spacing between the flat -top wires. M, the wire length from the outside to the center of each cross -over. D, the spacing lengthwise between sections. A (1/4), the approximate length for a quarter-wave stub. A (''s), the approximate length for a half -wave stub. A (3/4), the approximate length for a three -quarter wave stub. X, the approximate distance above the shorting wire of the stub for the connection of a 600 -ohm line. This distance, as given in the table, is approximately correct only for 2- section flat-tops. For single- section types it will be smaller and for 3- and 4- section types it will be larger. The lengths given for a half -wave stub are applicable only to single -section center -fed flat-tops. To be certain of sufficient stub length, it is advisable to make the stub a foot or so longer than shown in the table, especially with the end -fed types. The lengths, A, are measured from the point where the stub connects to the flat-top. Both the center and end -fed types may be used horizontally. However, where a vertical antenna is desired, the flat -tops can be turned on end. In this case, the end -fed types may be more convenient, feeding from the lower end. L www.americanradiohistory.com HANDBOOK Endlire Arrays Normally the antenna tank will be located as the transmitter, to facilitate adjustment when changing frequency. In this case it is recommended that the link coupled tank be located across the room from the transmitter if much power is used, in order to minimize r -f feedback difficulties which might occur as a result of the asymmetrical high impedance feed. If tuning of the antenna tank from the transmitter position is desired, flexible shafting can be run from the antenna tank condenser to a control knob at the transmitter. The lower end of the driven element is quite "hot" if much power is used, and the lead -in insulator should be chosen with this in mind. The ground connection need not have very low resistance, as the current flowing in the ground connection is comparatively small. A stake or pipe driven a few feet in the ground will suffice. However, the ground lead should be of heavy wire and preferably the length should 'not exceed about 10 feet at 7 Mc. or about 20 feet at 4 Mc. in order to minimize reactive effects due to its inductance. If it is impossible to obtain this short a ground lead, a piece of screen or metal sheet about four feet square may be placed parallel to the earth in a convenient location and used as an artificial ground. A fairly high C/L ratio ordinarily will be required in the antenna tank in order to obtain adequate coupling and loading. in the same room 23 -7 End -Fire Directivity By spacing two half-wave dipoles, or colinear arrays, at a distance of from 0.1 to 0.25 wavelength and driving the two 180° out of phase, directivity is obtained through the two wires at right angles to them. Hence, this type of bidirectional array is called end fire. A better idea of end -fire directivity can be obtained by referring to figure 10. Remember that end-fire refers to the radiation with respect to the two wires in the array rather than with respect to the array as a whole. The vertical directivity of an end -fire bidirectional array which is oriented horizontally can be increased by placing a similar end fire array a half wave below it, and excited in the same phase. Such an array is a combination broadside and end -fire affair. Flat -Top very effective bidirectional end -fire array is the Kraus or 8JK Hai-Top Beam. Essentially, this antenna consists of two closespaced dipoles or colinear arrays. Because of the close spacing, it is possible to obtain the Kraus Beam A 469 proper phase relationships in multi- section flat tops by crossing the wires at the voltage loops, rather than by resorting to phasing stubs. This greatly simplifies the array. (See figure 23.) Any number of sections may be used, though the one- and two -section arrangements are the most popular. Little extra gain is obtained by using more than four sections, and trouble from phase shift may appear. A center -fed single- section flat -top beam cut according to the table, can be used quite successfully on its second harmonic, the pattern being similar except that it is a little sharper. The single- section array can also be used on its fourth harmonic with some success, though there then will be four cloverleaf lobes, much the same as with a full -wave antenna. If a flat -top beam is to be used on more than one band, tuned feeders are necessary. The radiation resistance of a flat-top beam is rather low, especially when only one section is used. This means that the voltage will be high at the voltage loops. For this reason, especially good insulators should be used for best results in wet weather. The exact lengths for the radiating elements are not especially critical, because slight deviations from the correct lengths can be compensated in the stub or tuned feeders. Proper stub adjustment is covered in Chapter Twentyfive. Suitable radiator lengths and approximate stub dimensions are given in the accompanying design table. Figure 23 shows top views of eight types of flat -top beam antennas. The dimensions for using these antennas on different bands are given in the design table. The 7- and 28 -Mc. bands are divided into two parts, but the dimensions for either the low- or high -frequency ends of these bands will be satisfactory for use over the entire band. In any case, the antennas are tuned to the frequency used, by adjusting the shorting wire on the stub, or tuning the feeders, if no stub is used. The data in the table may be extended to other bands or frequencies by applying the proper factor. Thus, for 50 to 52 Mc. operation, the values for 28 to 29 Mc. are divided by 1.8. All of the antennas have a bidirectional horizontal pattern on their fundamental frequency. The maximum signal is broadside to the flat top. The single- section type has this pattern on both its fundamental frequency and second harmonic. The other types have four main lobes of radiation on the second and higher harmonics. The nominal gains of the different types over a half -wave comparison antenna are as follows: single- section, 4 db; two- section, 6 db; four- section, 8 db. The maximum spacings given make the beams less critical in their adjustments. Up www.americanradiohistory.com L L L RADIO THE High Frequency Directive Antennas 468 DI DI D2 3 saw END-LINK COIL TO TUNE FREQUENCY C. 100 LUF DIMENSIONS 30011 RIBBON LINE DIMENSIONS IOM. 134. 20M. GAIN APPROX. 7.5 Da IT' 22,3 33.6" tree 22.9- 34'6' L D Figure 21 THE "SIX- SHOOTER" BROADSIDE ARRAY wire line should be employed if the antenna is used with a high power transmitter. To tune the reflector, the back of the antenna is aimed at a nearby field -strength meter and the reflector stub capacitor is adjusted for minimum received signal at the operating frequency. This antenna provides high gain for its small size, and is recommended for 28 -Mc. work. The elements may be made of number 14 enamel wire, and the array may be built on a light bamboo or wood framework. The "Six- Shooter" Broadside Array As a good compromise between gain, directivity, compactness, mechanical simplicity, ease of adjustment, and band width the array of figure 21 is recommended for the 10 to 30 Mc. range when the additional array width and greater directivity are not obtainable. The free space directivity gain is approximately 7.5 db over one element, and the practical dx signal gain over one element at the same average elevation is of about the same magnitude when the array is sufficiently elevated. To show up to best advantage the array should be elevated sufficiently to put the lower elements well in the clear, and preferably at least 0.5 wavelength above ground. Another application of vertical orientation of the raBroadside Curtain diating elements of an array in order to obtain low angle radiation at the lower end of the h -f range with low pole heights is illustrated in figure 22. When precut to the specified dimensions this single pattern array will perform well over the 7 -Mc. amateur band or the 4 -Mc. amateur phone band. For the 4 -Mc. band the required two poles need be only 70 feet high, and the array will provide a practical signal The "Bobtail" Bidirectional 32 a COAXIAL LINE DI Dz 40M. 60M. 126, 33, 60, D3 30.7036 34'TOM' "BOBTAIL" Figure 22 BIDIRECTIONAL BROAD- SIDE CURTAIN FOR THE 7 -MC. OR THE 4.0 -MC. AMATEUR BANDS This simple vertically polarized array provides low angle radiation and response with comparatively low pole heights, and is very effective for dx work on the 7 -Mc. band or the 4.0 -Mc. phone band. Because of the phase relationships, radiation from the horizontal portion of the antenna is effectively suppressed. Very little current flows in the ground lead to the coupling tank; so an elaborate ground system is not required, and the length of the ground lead is not critical so long os it uses heavy wire and is reasonably short. gain averaging from 7 to 10 db over a horizontal half-wave dipole utilizing the same pole height when the path length exceeds 2500 miles. The horizontal directivity is only moderate, the beam width at the half power points being slightly greater than that obtained from three cophased vertical radiators fed with equal currents. This is explained by the fact that the current in each of the two outer radiators of this array carries only about half as much current as the center, driven element. While this "binomial" current distribution suppresses the end -fire lobe that occurs when an odd number of parallel radiators with half-wave spacing are fed equal currents, the array still exhibits some high -angle radiation and response off the ends as a result of imperfect cancellation in the flat top portion. This is not sufficient to affect the power gain appreciably, but does degrade the discrimination somewhat. A moderate amount of sag can be tolerated at the center of the flat top, where it connects to the driven vertical element. The poles and antenna tank should be so located with respect to each other that the driven vertical element drops approximately straight down from the flat top. www.americanradiohistory.com HANDBOOK Broadside Arrays 467 EACN SIOE CDR CAIN sT EACH SIDE REFLECTOR C' LLF PUNK FOR M MIN PICKUP OF BEI M. NOTE I I RM REAR 'RADIATOR 5% 8.107 Id SPACED s,oe LENCTN- rI' B IT T )^ ON BALANCED CED LINE TUNING UNIT OR TRANSMITTER. FOR 2I MC FOR 74 MC. ELEMENT SPACING PB- FOR LACS BAND. STUD LENCT PPROX IS' 20 DIMENSIONS 101.4 ISM. FOR 21 MC. FOR 14 MC. 50M. B'B 2YY SS' D B'S 12.57 O SECTION 151E. WIRE SPACED 4- GAIN APPROX. DB 150.11 LINE TO TRANSMITTER THE Figure THE Figure 20 CUBICAL -QUAD ANTENNA FOR THE 10 -METER BAND 19 "BI- SQUARE" BROADSIDE ARRAY This bidirectional array Is related to the and in spite of the oblique elements, is horizontally polarized. It has slightly less gain and directivity than the Lazy H, the free space directivity gain being approximately 4 db. Its chief advantage is the fact that only a single pole is required for "Lazy H," support, and two such arrays may be supported from a single pole without interaction if the planes of the elements are at right angles. A 600 -ohm line may be substituted for the Twin-Lead, and either operated os a resonant line, or made non -resonant by the incorporation of a matching stub. still worthwhile, being approximately 4 db over a half-wave horizontal dipole at the same average elevation. When two Bi -Square arrays are suspended at right angles to each other (for general coverage) from a single pole, the Q sections should be well separated or else symmetrically arranged in the form of a square (the diagonal conductors forming one Q section) in order to minimize coupling between them. The same applies to the line if open construction is used instead of Twin -Lead, but if Twin -Lead is used the coupling can be made negligible simply by separating the two Twin -Lead lines by at least two inches and twisting one TwinLead so as to effect a transposition every foot or so. When tuned feeders are employed, the BiSquare array can be used on half frequency as an end -fire vertically polarized array, giving a slight practical dx signal gain over a vertical half -wave dipole at the same height. A second Bi -Square serving as a reflector may be placed 0.15 wavelength behind this antenna to provide an overall gain of 8.5 db. The reflector may be tuned by means of a quarterwave stub which has a moveable shorting bar at the bottom end. The stub is used as a substitute for the Q- section, since the reflector employs no feed line. smaller version of the BiSquare antenna is the Cubical -Quad antenna. Two halfwaves of wire are folded into a square that is one -quarter wavelength on a side, as shown in figure 20. The arraÿ radiates a horizontally polarized signal. A reflector placed 0.15 wavelength behind the antenna provides an overall gain of some 6 db. A shorted stub with a paralleled tuning capacitor is used to resonate the The "CubicalQuad" Antenna A reflector. The Cubical -Quad is fed with a 150-ohm line, and should employ some sort of antenna tuner at the transmitter end of the line if a pinetwork type transmitter is used. There is a small standing wave on the line, and an open www.americanradiohistory.com THE High Frequency Directive Antennas 466 D L + RADIO ' L L - L ' 1111111111 GAIN APPROX. 5 DB 9004 LINE DIMENSION L 10M 13'9- DIMENSIONS 10M. GAIN APPRO*. 3 DB I5M. 20M. L 13'3- 22' 32'10. S 20' 30' 40' P 14.2 21'3 213'4 D 3'Y 7.0- X -ARRAY 7511 TRANSMISSION LINE FOR 17 28 MC., 21 17.3 ANTENNA TUNER TRANSMITTER Figure 18 DOUBLE -BRUCE ARRAY FOR 10, 15, AND 20 METERS If a 600 -ohm feed line is used, the 20 -meter array will also perform on 10 meters as o Sterba curtain, with an approximate gain of 9 db. MC., OR 14 MC. The entire array (with the exception of the 75 -ohm feed line) is constructed of 300 -ohm ribbon line. Be sure phasing lines (P) are poled correctly, as shown. in a vertical plane and properly phased, a simplified form of in -phase curtain is formed, providing an overall gain of about 6 db. Such an array is shown in figure 17. In this X- array, the four dipoles are all in phase, and are fed by four sections of 300 -ohm line, each one half wavelength long, the free ends of all four lines being connected in parallel. The feed impedance at the junction of these four lines is about 75 ohms, and a length of 75 -ohm Twin -Lead may be used for the feedline to the array. An array of this type is quite small for the 28 -Mc. band, and is not out of the question for the 21 -Mc. band. For best results, the bottom section of the array should be one -half wavelength above ground. The Double -Bruce Array 20 /IOM. THE Figure THE I5M. 113' The Bruce Beam consists of a long wire folded so that vertical elements carry in -phase currents while the horizontal elements carry out of phase currents. Radiation from the horizontal sections is low since only a small current flows in this part of the wire, and it is largely phased-out. Since the height of the Bruce Beam is only one -quarter wavelength, the gain per linear foot of array is quite low. Two Bruce Beams may be combined as shown in figure 18 to produce the Double Bruce array. A four section Double Bruce will give a vertically polarized emission, with a power gain of 5 db over a simple dipole, and is a very simple beam to construct. This antenna, like other so- called "broadside" arrays, radiates maximum power at right angles to the plane of the array. The feed impedance of the Double Bruce is about 750 ohms. The array may be fed with a one -quarter wave stub made of 300 -ohm ribbon line and a feedline made of 150 -ohm ribbon line. Alternatively, the array may be fed directly with a wide- spaced 600 -ohm transmission line (figure 18). The feedline should be brought away from the Double Bruce for a short distance before it drops downward, to prevent interaction between the feedline and the lower part of the center phasing section of the array. For best results, the bottom sections of the array should be one -half wavelength above ground. Arrays such as the X -array and the Double Bruce are essentially high impedance devices, and exhibit relatively broad -band characteristics. They are less critical of adjustment than a parasitic array, and they work well over a wide frequency range such as is encountered on the 28 -29.7 Mc. band. The "Bi- Square" Broadside Array Illustrated in figure 19 is a simple method of feeding a small broadside array first described by W6BCX several years ago as a practical method of suspending an effective array from a single pole. As two arrays of this type can be supported at right angles from a single pole without interaction, it offers a solution to the problem of suspending two arrays in a restricted space with a minimum of erection work. The free space directivity gain is slightly less than that of a Lazy I1, but is www.americanradiohistory.com HANDBOOK Broadside Arrays 465 NON - RESONANT FELDER GAIN APPROX. 6 DB GAIN APPROX. 8 DB Figure 16 THE STERBA CURTAIN ARRAY Approximate directive gains along with alternative feed methods are shown. GAIN APPROX. 8 DB sent points of maximum current. All arrows should point in the same direction in each portion of the radiating sections of an antenna in order to provide a field in phase for broadside radiation. This condition is satisfied for the arrays illustrated in figure 16. Figures 16A and 16C show simple methods of feeding a short Sterba curtain, while an alternative method of feed is shown in the higher gain antenna of figure 16B. In the case of each of the arrays of figure 16, and also the "Lazy H" of figure 15, the array may be made unidirectional and the gain increased by 3 db if an exactly similar array is constructed and placed approximately 14 wave behind the driven array. A screen or mesh of wires slightly greater in area than the antenna array may be used instead of an additional array as a reflector to obtain a unidirectional system. The spacing between the reflecting wires may vary from 0.05 to 0.1 wavelength with the spacing between the reflecting wires the smallest directly behind the driven elements. The wires in the untuned reflecting system should be parallel to the radiating elements of the array, and the spacing of the complete reflector system should be approximately 0.2 to 0.25 wavelength behind the driven elements. On frequencies below perhaps 100 Mc. it normally will be impracticable to use a wire screen reflector behind an antenna array such as a Sterba curtain or a "Lazy H." Parasitic elements may be used as reflectors or directors, but parasitic elements have the disadvantage that their operation is selective with respect to relatively small changes in frequency. Nevertheless, parasitic reflectors for such arrays are quite widely used. In section 23 -5 it was shown how two dipoles may be arranged in phase to provide a power gain of (some) 3 db. If two such pairs of dipoles are stacked The X -Array LAZY -H AND STERBA (STACKED DIPOLE) DESIGN TABLE FREQUENCY IN MC. 7.0 7.3 14.0 14.2 14.4 21.0 21.5 27.3 28.0 29.0 50.0 52.0 54.0 144.0 146.0 148.0 www.americanradiohistory.com L, 68'2" 65'10" 34'1" 33'8" 33'4" 22'9" 22'3" 17'7" 17' 16'6" 9'7" 9'3" 8'10" 39.8" 39" 38.4" L. 70' 67'6" 35' 34.7" 34'2" 23'3" 22'9" 17'10" 17'7" 17' 9'10" 9'S" 9'1" 40.5" 40" 39.5" L, 35' 33'9" 17'6" 17'3" 17' 11'B" 11'5" 8'11" 8'9" 8'6" 4'11" 4'8" 4'6" 20.3' 20" 19.8" 464 THE High Frequency Directive Antennas of a colinear antenna is proportional to the RADIO Li LI overall length, whether the individual radiating elements are 1/4 wave, 1/2 wave or 1/4 wave in length. The gain of two colinear half waves may be increased by increasing the physical spacing between the elements, up to a maximum of about one half wavelength. If the half wave elements are fed with equal lengths of transmission line, poled correctly, a gain of about 3.3 db is produced. Such an antenna is shown in figure 13. By means of a phase reversing switch, the two elements may be operated out of phase, producing a cloverleaf pattern with slightly less maximum gain. A three element "precut" array for 40 meter operation is shown in figure 14. It is fed directly with 300 ohm "ribbon line," and may be matched to a 52 ohm coaxial output transmitter by means of a Balun, such as the Barker & illiamson 3975. The antenna has a gain of about 3.2 db, and a beam width at half -power points of 40 degrees. Spaced Half Wave Antennas 23 -6 OUARTER-WAVE STUR NON -RESONANT FEED LIN CAIN APPROX. 5.5 DR Broadside Arrays Colinear elements may be stacked above or below another string of colinear elements to produce what is commonly called a broadside array. Such an array, when horizontal elements are used, possesses vertical directivity in proportion to the number of broadsided (vertically stacked) sections which have been used. Since broadside arrays do have good vertical directivity their use is recommended on the 14 -Mc. band and on those higher in frequency. One of the most popular of simple broadside arrays is the "Lazy 11" array of figure 15. Horizontal colinear elements stacked two above two make up this antenna system which is highly recommended for work on frequencies above perhaps 14 -Mc. when moderate gain without too much directivity is desired. It has high radiation resistance and a gain of approximately 5.5 db. The high radiation resistance results in low voltages and a broad resonance curve, which permits use of inexpensive insulators and enables the array to be used over a fairly wide range in frequency. For dimensions, see the stacked dipole design table. Vertical stacking may be applied Dipoles to strings of colinear elements longer than two half waves. In such arrays, the end quarter wave of each string of radiators usually is bent in to meet Stacked 2 RESONANT FEED LINE Figure 15 THE "LAZY H" ANTENNA SYSTEM Stacking the colinear pairs gives both horizontal and vertical directivity. As shown, the array will give about 5.5 db gain. Note that the array may be fed either at the center of the phasing section or at the bottom; if fed at the bottom the phasing section must be twisted through 180 °. a similar bent quarter wave from the opposite end radiator. This provides better balance and better coupling between the upper and lower elements when the array is current -fed. Arrays of this type are shown in figure 16, and are commonly known as curtain arrays. Correct length for the elements and stubs can be determined for any stacked dipole array from the Stacked -Dipole Design Table. In the sketches of figure 16 the arrowheads represent the direction of current flow at any given instant. The dots on the radiators repre- www.americanradiohistory.com Colinear Arrays HANDBOOK COLINEAR ANTENNA DESIGN CHART FREQUENCY IN MC. Li La 16'8' 22'e" 14.2 .0 L3 e'6 17' 1+'e 33'e 23'3 34.7 e7' 66'6" 34'4 61'6 68'2 120' 133' 3.e RuCI FMC) s 26.5 21.2 7.15 463 F(I) A B A-B =15011 FEED POINT GAINAPPROX. 3D6 17'3 123' 136'5- Figure 12 DOUBLE EXTENDED ZEPP ANTENNA For best results, antenna should be tuned to operating frequency by means of griddip oscillator. simple colinear antenna array very effective radiating system for the 3.5 -Mc. and 7.0 -Mc. bands, but its use is not recommended on higher frequencies since such arrays do not possess any vertical directivity. The elevation radiation pattern for such an array is essentially the same as for a half-wave dipole. This consideration applies whether the elements are of normal length or are extended. The colinear antenna consists of two or more radiating sections from 0.5 to 0.65 wavelengths long, with the current in phase in each section. The necessary phase reversal between sections is obtained through the use of resonant tuning stubs as illustrated in figure 11. The gain of a colinear array using half -wave elements (in decibels) is approximately equal to the number of elements in the array. The exact figures are as follows: 2 3 4 6 Number of Elements 5 Gain in Decibels 1.8 3.3 4.5 5.3 6.2 As additional in -phase colinear elements are added to a doublet, the radiation resistance goes up much faster than when additional half waves are added out of phase (harmonic operated antenna). For a colinear array of from 2 to 6 elements, Colinear Arrays The is h~- e'(,`1 a i' F(1AC) the terminal radiation resistance in ohms at any current loop is approximately 100 times the number of elements. It should be borne in mind that the gain from a colinear antenna depends upon the sharpness of the horizontal directivity since no vertical directivity is provided. An array with several colinear elements will give considerable gain, but will have a sharp horizontal radiation pattern. The gain of a conventional two- element Franklin colinear antenna can be increased to a value approaching that obtained from a three -element Franklin, simply by making the two radiating elements 230° long instead of 180° long. The phasing stub is shortened correspondingly to maintain the whole array in resonance. Thus, instead of having 0.5 -wavelength elements and 0.25-wavelength stub, the elements are made 0.64 wavelength long and the s tub is approximately 0.11 wavelength Double Extended Zepp long. Dimensions for the double extended Zepp are given in figure 12. The vertical directivity of a colinear antenna having 230° elements is the same as for one having 180° elements. There is little advantage in using extended sections when the total length of the array is to be greater than about 1.5 wavelength overall since the gain r'-- 65 e -Ai MID 32'9 PHASE -REVERSING SWITCH FOR CLOVERLEAF PATTERN MAKE STUBS 65 6 -H TOeE. 14 65 e-+{ e ms 32'9' - OFl WIRE,SPACED - "-3OOR RIBBON TO TRANSMITTER, ANY LENGTH GAIN APPROX. 3 DB Figure 13 Figure 14 TWO COLINEAR HALF -WAVE ANTENNAS IN PHASE PRODUCE A 3 DB GAIN WHEN PRE -CUT LINEAR ARRAY FOR 40 -METER SEPARATED ONE -HALF WAVELENGTH OPERATION www.americanradiohistory.com High Frequency Directive Antennas 462 ---Lt f 4- L2 THE RADIO L2 Lt -----e-- - _ - L! PLANE OF WIRES END VIEW L3 QUARTER-WAVE STUBS NON- RESONANT FEED LIN 14) 5= = ISO. OUT OF PHASE (FLAT -TOP BEAM, ETC.) I / IN PHASE (LAZY H, SIERRA CURTAIN) Figure 10 RADIATION PATTERNS OF A PAIR OF DIPOLES OPERATING WITH IN -PHASE EXCITATION, AND WITH EXCITATION GAIN APPROS Figure 4 S DB 11 THE FRANKLIN OR COLINEAR ANTENNA ARRAY An antenna of this type, regardless of the number of elements, attains all of its directivity through sharpening of the horizontal or azimuth radiation pattern; no vertical directivity is provided. Hence a long antenna of this type has an extremely sharp azimuth pattern, but no vertical directivity. 180° OUT OF PHASE If the dipoles are oriented horizontally most of the directivity will be in the vertical plane; if they are oriented vertically most of the directivity will be in the horizontal plane. and 180° (45 °, 90 °, and 135° for instance), the pattern is unsymmetrical, the radiation be- ing greater in one direction than in the oppo- site direction. With spacings of more than 0.8 wavelength, more than two main lobes appear for all phasing combinations; hence, such spacings are seldom used. With the dipoles driven so as to be in phase, the most effective spacing is between 0.5 and 0.7 wavelength. The latter provides greater gain, but minor lobes are present which do not appear at 0.5- wavelength spacing. The radiation is broadside to the plane of the wires, and the gain is slightly greater than can be obtained from two dipoles out of phase. The gain falls off rapidly for spacings less than 0.375 wavelength, and there is little point in using spacing of 0.25 wavelength or less with in -phase dipoles, except where it is desirable to increase the radiation resistance. (See Multi In -Phase Spacing Wire Doublet.) dipoles are fed 180° out of phase, the directivity is through the plane of the wires, and is greatest with close spacing, though there is but difference in the pattern after the spacing is made less than 0.125 wavelength. The radiation resistance becomes so low for spacings of less than 0.1 wavelength that such spacings are not practicable. Out of Phase Spacing When the little In the three foregoing examples, most of the a plane at a right angle to the wires, though when out of phase, the directivity is in a line through the wires, and when in phase, the directivity is broadside to them. Thus, if the wires are oriented verti- directivity provided is in cally, mostly horizontal directivity will be provided. If the wires are oriented horizontally, most of the directivity obtained will be vertical directivity. To increase the sharpness of the directivity in all planes that include one of the wires, additional identical elements are added in the line of the wires, and fed so as to be in phase. The familiar H array is one array utilizing both types of directivity in the manner prescribed. The two -section Kraus flat -top beam is another. These two antennas in their various forms are directional in a horizontal plane, in addition to being low -angle radiators, and are perhaps the most practicable of the bidirectional stacked -dipole arrays for amateur use. More phased elements can be used to provide greater directivity in planes including one of the radiating elements. The fl then becomes a Sterba- curtain array. For unidirectional work the most practicable stacked -dipole arrays for amateur -band use are parasitically- excited systems using relatively close spacing between the reflectors and the directors. Antennas of this type are described in detail in a later chapter. The next most practicable unidirectional array is an H or a Sterba curtain with a similar system placed approximately one -quarter wave behind. The use of a reflector system in conjunction with any type of stacked -dipole broadside array will increase the gain by 3 db. www.americanradiohistory.com HANDBOOK Antenna Rhombic The 461 J, Figure 8 TYPICAL RHOMBIC ANTENNA DESIGN The antenna system illusabove may be used over the frequency ronge from 7 to 29 Mc. without change. The directivity of the system may be reversed by the system discussed in the text. trated LINE TO TX N14 SPACED e' S. 214 FEET SPACING BETWEEN SIDES TOTAL LENGTH TERMINATING LINE OF 250' OF N 26 NICHROME SPACED 6" AND B00 -OHM 16 -WATT 5112 FEET H50 This antenna will give about 11 db gain in the 14.0 -Mc. band. The approximate gain of a rhombic antenna over a dipole, both above normal soil, is given in figure 9. A considerable amount of directivity is lost when the terminating resistor is left off the end and the system is operated as a resonant antenna. If it is desired to reverse the direction of the antenna it is much better practice to run transmission lines to both ends of the antenna, and then run the terminating line to the operating position. Then with the aid of two d -p -d -t switches it will be possible to connect either feeder to the antenna changeover switch and the other feeder to the terminating line, thus reversing the direction of the array and maintaining the same termination for bands. Stacked -Dipole 23 -5 Arrays The characteristics of a half -wave dipole already have been described. When another dipole is placed in the vicinity and excited either direction of operation. Figure 7 gives curves for optimum- design rhombic antennas by both the maximum-output method and the alignment method. The alignment method is about 1.5 db down from the maximum output method but requires only about 0.74 as much leg length. The height and tilt angle is the same in either case. Figure 8 gives construction data for a recommended rhombic antenna for the 7.0 through 29.7 Mc. either directly or parasitically, the resultant radiation pattern will depend upon the spacing and phase differential, as well as the relative magnitude of the currents. With spacings less than 0.65 wavelength, the radiation is mainly broadside to the two wires (bidirectional) when there is no phase difference, and through the wires (end fire) when the wires are 180° out of phase. With phase differences between 0° ILI J ie 215 p 14 w Figure 9 RHOMBIC ANTENNA GAIN Showing the theoretical gain of a rhombic antenna, in terms of the side length, over a half -wave antenna mounted at the same height above the same type of soil. CARBON RESISTOR AT 6 2-WATT 100-OHM END RESISTORS IN SERIES 1 312 LL 11 3 CO e z Z 3 .. .r...._... ..... .......... ..01......... ......... / . II N........ M. .. III J1 = CC w Ó .. ../NM ..... ........1,,.. .......,........... ....MM............ .... .............. ...........mm%_ WI . ........ .................. 30 2 3 4 Il "LENGTH 5 6 7 6 S 10 11 R 13 14 15 le Ti 16 /f 20 OF EACH LEG OF RHOMBIC IN WAVELENGTHS www.americanradiohistory.com 460 23-4 The Rhombic Antenna .... .... Rhombic Termination H RADIO c..,ww.cwa a no .a.n u .. .. .. . . The terminated rhombic or diamond is probably the most effective directional antenna that is practical for amateur communication. This antenna is non -resonant, with the result that it can be used on three amateur bands, such as 10, 20, and 40 meters. When the antenna is non -resonant, i.e., properly terminated, the system is undirectional, and the wire dimensions are not critical. . . . . o is terminated resistance of a value When the free end with a between 700 and 800 ohms the backwave is eliminated, the forward gain is increased, and the antenna can be used on several bands without changes. The terminating resistance should be capable of dissipating one -third the power output of the transmitter, and should have very little reactance. For medium or low power transmitters, the non -inductive plaque resistors will serve as a satisfactory termination. Several manufacturers offer special resistors suitable for terminating a rhombic antenna. The terminating device should, for technical reasons, present a small amount of inductive reactance at the point of termination. A compromise terminating device commonly used consists of a terminated 250 -foot or longer length of line, made of resistance wire which does not have too much resistance per unit length. If the latter qualification is not met, the reactance of the line will be excessive. A 250 -foot line consisting of no. 25 nichrome wire, spaced 6 inches and terminated with 800 ohms, will serve satisfactorily. Because of the attenuation of the line, the lumped resistance at the end of the line need dissipate but a few watts even when high power is used. A half-dozen 5000 -ohm 2 -watt carbon resistors in parallel will serve for all except very high power. The attenuating line may be folded back on itself to take up less room. The determination of the best value of terminating resistor may be made while receiving, if the input impedance of the receiver is approximately 800 ohms. The value of resistor which gives the best directivity on reception will not give the most gain when transmitting, but there will be little difference between the two conditions. The input resistance of the rhombic which is reflected into the transmission line that feeds it is always somewhat less than the terminating resistance, and is around 700 to 750 ohms when the terminating resistor is 800 ohms. THE High Frequency Directive Antennas r r M Ir n. WAVE ANGLE Figure A sr ar Zr 7 RHOMBIC ANTENNA DESIGN TABLE Design data is given in terms of the wave angle (vertical angle of transmission and reception) of the antenna. The lengths I are for the "maximum output" design; the shorter are for the "alignment" method lengths which gives approximately 1.5 db less gain with o considerable reduction in the space required for the antenna. The values of side length, tilt angle, and height for a given wave angle are obtained by drawing o vertical line upward from the desired wave angle. I' The antenna should be fed with a non -resonant line having a characteristic impedance of 650 to 700 ohms. The four corners of the rhombic should be at least one -half wavelength above ground for the lowest frequency of operation. For three-band operation the proper tilt angle ,;4 for the center band should be observed. The rhombic antenna transmits tally- polarized wave at a a horizon- relatively low angle above the horizon. The angle of radiation (wave angle) decreases as the height above ground is increased in the same manner as with a dipole antenna. The rhombic should not be tilted in any plane. In other words, the poles should all be of the same height and the plane of the antenna should be parallel with the ground. www.americanradiohistory.com HANDBOOK I/l i 6/ L / M . I If 1:/ 1P.i >lo11SL The Antenna V 459 13 3 w. ' b TRAN3MIT RECEIVE t yl o Jul u p Figure TYPICAL "V" Z z 5 BEAM ANTENNA M16 411(Ii 7 3=1111/411011111 2 RIMIIIIIME111101111111.11MMIEll oo 2 2 4 3 6 LENGTH OF SIDE 7 "L" other removes two of the four main lobes, and increases the other two in such a way as to form two lobes of still greater magnitude. The correct wire lengths and the degree of the angle b are listed in the V- Antenna Design Table for various frequencies in the 10 -, 20and 40 -meter amateur bands. Apex angles for all side lengths are given in figure 4. The gain of a "V" beam in terms of the side length when optimum apex angle is used is given in figure 6. The legs of a very long V antenna are usually so arranged that the included angle is twice the angle of the major lobe from a single wire if used alone. This arrangement concentrates the radiation of each wire along the bisector of the angle, and permits part of the other lobes to cancel each other. üith legs shorter than 3 wavelengths, the best directivity and gain are obtained with a somewhat smaller angle than that determined by the lobes. Optimum directivity for a one wave V is obtained when the angle is 90° rather than 180 °, as determined by the ground pattern alone. If very long wires are used in the V, the angle between the wires is almost unchanged when the length of the wires in wavelengths is altered. However, an error of a few degrees causes a much larger loss in directivity and gain in the case of the longer V than in the shorter one. The vertical angle at which the wave is best transmitted or received from a horizontal V antenna depends largely upon the included angle. The sides of the V antenna should be at least a half wavelength above ground; commercial practice dictates a height of approximately a full wavelength above ground. =2r 6° 70 L 6'so 26000 29000 34'6" 69'6 33.6 21100 L =4a L =BT 6 =52 6F39 260' 67.3" 140' 135' 45'9" 91.9" 163' 21300 45'4" 91'4" 162'6 366' 365' 14050 14150 14250 69' 7020 7100 7200 66'6" 66'2' 136'2' 136.6' 134.10" 12 ground, in terms of the side length L. V- ANTENNA DESIGN TABLE L= ñ 11 Figure 6 DIRECTIVE GAIN OF A "V" BEAM This curve shows the approximate directive gain of a V beam with respect to a half-wave antenna located the same distance above for a long wire. The reaction of one upon the FREQUENCY IN KILOCYCLES 10 271' 139' 136' 137' 279' 277' 275' 356' 555' 276' 275' 271' 556' 1120' 1106' 1060' www.americanradiohistory.com 552' 545' 552' High Frequency Directive Antennas 458 THE RADIO 4A 4ZA LONG- ANTENNA DESIGN TABLE APPROXIMATE LENGTH IN FEET-END-FED ANTENNAS FREQUENCY IN M IZA 1A 3A 84 67 101 118 104 136 1/2 91 3/4 114 1/2 114 3/4 92 115 137 171 29 26 50 52 67 69 21.4 21.2 21.0 45 45 1M 66 66 1/4 911/2 66 1/2 14.2 14.0 67 1/2 88 1/2 7.3 7.15 7.0 .0 3.6 3.6 3.5 2.0 1.9 1.6 451/2 102 2 I39 103 1/2 174 138 136 1/2 137 206 276 207 207 1/2 277 277 1/2 346 347 348 240 232 362 465 616 361 511 60 268 403 414 540 555 676 696 1230 1280 274 480 304 725 972 763 1020 532 605 1060 152 122 135 140 160 1/2 160 3.4 165 1/2 163 3/4 2091/2 137 161 166 206 209 240 244 275 279 310 418 66 555 625 557 627 416 467 488 356 628 633 977 1030 1090 1120 1100 1160 136 3.41 730 770 612 633 900 950 977 157 209 3/4 210 31 1220 1473 form of a V, it is possible to make two of the maximum lobes of one leg shoot in the same direction as two of the maximum lobes of the other leg of the V. The resulting antenna is bidirectional (two opposite directions) for the main lobes of radiation. Each side of the V can be made any odd or even number of quarter wavelengths, depending on the method of feeding the apex of the V. The complete system must be a multiple of half waves. If each leg is an even number of quarter waves long, the antenna must be voltage -fed at the apex; if an odd number of quarter waves long, current feed must be used. By choosing the proper apex angle, figure and figure 5, the lobes of radiation from the two long -wire antennas aid each other to form a bidirectional beam. Each wire by itself would have a radiation pattern similar to that 4 The V Antenna If 3 17 One of the most practical methods of feeding a long -wire antenna is to bring one end of it into the radio room for direct connection to a tuned antenna circuit which is link- coupled through a harmonic- attenuating filter to the transmitter. The antenna can be tuned effectively to resonance for operation on any harmonic by means of the tuned circuit which is connected to the end of the antenna. A ground is sometimes connected to the center of the tuned coil. If desired, the antenna can be opened and current -fed at a point of maximum current b' means cf low- impedance ribbon line, or by a quarter -wave matching section and open line. 23 -3 X +A 2X 33 34 two long -wire antennas are built in the ISO 140 Figure 4 INCLUDED ANGLE FOR A 120 "V" BEAM Showing the included angle between the legs of a V beam for various leg lengths. For optimum alignment of the radiation lobe at the correct vertical angle with leg lengths less thon three wavelengths, the optimum Included angle is shown by the dashed curve. 40 20 o o 4 LENGTH IN 10 6 "L' 12 WAVELENGTHS www.americanradiohistory.com HANDBOOK Long Wire Radiators 457 LONG STRAIGHT WIRE ANTENNAS Figure 3 DIRECTIVE GAIN OF LONG -WIRE ANTENNAS 2 °o 2 3 4 3 7 e s 10 DB POWER RATIO OF MAIN LOBE TO A DIPOLE Types of There is an enormous variety of directive antenna arrays that can give a substantial power gain in the desired direction of transmission or reception. However, some are more effective than others requiring the same space. In general it may be stated that long -wire antennas of various types, such as the single long wire, the V beam, and the rhombic, are less effective for a given space than arrays composed of resonant elements, but the long wire arrays have the significant advantage that they may be used over a relatively large frequency range while resonant arrays are usable only over a quite narrow frequency band. The horizontal radiation pattern of such antennas depends upon the vertical angle of radiation being considered. If the wire is more than 4 wavelengths long, the maximum radiation at vertical angles of 15° to 20° (useful for dx) is in line with the wire, being slightly greater a few degrees either side of the wire than directly off the ends. The directivity of the main lobes of radiation is not particularly sharp, and the minor lobes fill in between the main lobes to permit working stations in nearly all directions, though the power radiated broadside to the radiator will not be great if the radiator is more than a few wavelengths long. The directive gain of long -wire antennas, in terms of the wire length in wavelengths is given in figure 3. Long Wire Radiators To maintain the out -of-phase condition in adjoining half -wave elements throughout the length of the radiator, it is necessary that a harmonic antenna be fed either at one end or at a current loop. If fed at a voltage loop, the adjacent sections will be fed in phase, and a different radiation pattern will result. The directivity of a long wire does not increase very much as the length is increased beyond about 15 wavelengths. This is due to the fact that all long -wire antennas are adversely affected by the r -f resistance of the wire, and because the current amplitude begins to become unequal at different current loops, as a result of attenuation along the wire caused by radiation and losses. As the length is increased, the tuning of the antenna becomes quite broad. In fact, a long wire about 15 waves long is practically aperiodic, and works almost equally well over a wide range of frequencies. Directive Arrays 23 -2 Harmonically operated long wires radiate better in certain directions than others, but cannot be considered as having appreciable directivity unless several wavelengths long. The current in adjoining half -wave elements flows in opposite directions at any instant, and thus, the radiation from the various elements adds in certain directions and cancels in others. A half -wave do u b l e t in free space has a "doughnut" of radiation surrounding it. A full wave has 2 lobes, 3 half waves 3, etc. When the radiator is made more than 4 half wavelengths long, the end lobes (cones of radiation) begin to show noticeable power gain over a half-wave doublet, while the broadside lobes get smaller and smaller in amplitude, even though numerous (figure 2). www.americanradiohistory.com 456 High Frequency Directive Antennas 0. M11Ì11111 7, ,,. 111111 ,°. U1111III ,O. S0. 20 ,O n. 1111111 - 1111 DOUBLE HOP . h. SINGLE HOP 1111111 30 SO 100 1111 11111 111111 1101 111111 11111 __ _eÌ 1111 111111 HID 1111h 1111.,11 1 1 1 300 500 UMW ,000 3000 ro 000 GREAT CIRCLE DISTANCE IN MILES Figure 1 OPTIMUM ANGLE OF RADIATION WITH RESPECT TO DISTANCES Shown above is o plot of the optimum angle of radiation for one -hop and two -hop communication. An operating frequency close to the optimum working frequency for the communication distance is assumed. a directive antenna than to increase transmitter power, if more than a few watts of power is being used. Directive antennas for the high- frequency range have been designed and used commercially with gains as high as 23 db over a simple dipole radiator. Gains as high as 35 db are common in direct -ray microwave communication and radar systems. A gain of 23 db represents a power gain of 200 times and a gain of 35 db represents a power gain of almost 3500 times. However, an antenna with a gain of only 15 to 20 db is so sharp in its radiation pattern that it is usable to full advantage only for point to -point work. The increase in radiated power in the desired direction is obtained at the expense of radiation in the undesired directions. Power gains of 3 to 12 db seem to be most practicable for amateur communication, since the width of a beam with this order of power gain is wide enough to sweep a fairly large area. Gains of 3 to 12 db represent effective transmitter power increases from 2 to 16 times. use There is a certain optimum vertical angle of radiation for sky -wave communication, this angle being dependent upon distance, frequency, time of day, etc. Energy radiated at an angle much lower than this optimum angle is largely lost, while radiation at angles much Horizontal Pattern vs. Vertical Angle 'l j THE RADIO ir ..®:4 1p 1 I ,., W1111'14 ,.',, '.I.iiats 1í ÿ vv 1 I q `,. `'1 , ..I -- et* ihoir HALT WAVE ANT. --711G.++..-, FULL WAVE ANT. 2 WAVES ANT. HORIZONTAL ANTENNAS IN FREE SPACE Figure 2 FREE -SPACE FIELD PATTERNS OF LONG -WIRE ANTENNAS The presence of the earth distorts the field pattern in such a manner that the azimuth pattern becomes a function of the elevation angle. higher than this optimum angle oftentimes is not nearly so effective. For this reason, the horizontal directivity pattern as measured on the ground is of no import when dealing with frequencies and distances dependent upon sky -wave propagation. It is the horizontal directivity (or gain or discrimination) measured at the most useful vertical angles of radiation that is of consequence. The horizontal radiation pattern, as measured on the ground, is considerably different from the pattern obtained at a vertical angle of 15 °, and still more different from a pattern obtained at a vertical angle of 30 °. In general, the energy which is radiated at angles higher than approximately 30° above the earth is effective at any frequency only for local work. For operation at frequencies in the vicinity of 14 Mc., the most effective angle of radiation is usually about 15° above the horizon, from any kind of antenna. The most effective angles for 10 -meter operation are those in the vicinity of 10 °. Figure 1 is a chart giving the optimum vertical angle of radiation for sky -wave propagation in terms of the great -circle distance between the transmitting and receiving antennas. www.americanradiohistory.com CHAPTER TWENTY -THREE High Frequency Antenna Arrays It is becoming of increasing importance in most types of radio communication to be capable of concentrating the radiated signal from the transmitter in a certain desired direction and to be able to discriminate at the receiver against reception from directions other than the desired one. Such capabilities involve the use of directive antenna arrays. Few simple antennas, except the single vertical element, radiate energy equally well in all azimuth (horizontal or compass) directions. justed. They all are dipoles, and the feeder system, if it does not radiate in itself, will have no effect on the radiation pattern. 23 -1 Directive Antennas When a multiplicity of radiating elements is located and phased so as to reinforce the radiation in certain desired directions and to neutralize radiation in other directions, a directive antenna array is formed. The function of a directive antenna when used for transmitting is to give an increase in signal strength in some direction at the expense of radiation in other directions. For reception, one might find useful an antenna giving little or no gain in the direction from which it is desired to receive signals if the antenna is able to discriminate against interfering signals and static arriving from other directions. A good directive transmitting antenna, however, can also be used to good advantage for reception. If radiation can be confined to a narrow beam, the signal intensity can be increased a great many times in the desired direction of transmission. This is equivalent to increasing the power output of the transmitter. On the higher frequencies, it is more economical to All horizontal antennas, except those specifically designed to give an omnidirectional azimuth radiation pattern such as the turnstile, have some directive properties. These properties depend upon the length of the antenna in wavelengths, the height above ground, and the slope of the radiator. The various forms of the half -wave horizontal antenna produce maximum radiation at right angles to the wire, but the directional effect is not great. Nearby objects also minimize the directivity of a dipole radiator, so that it hardly seems worth while to go to the trouble to rotate a simple half -wave dipole in an attempt to improve transmission and reception in any direction. The half -wave doublet, folded dipole, zepp, single- wire -fed, matched impedance, and Johnson Q antennas all have practically the same radiation pattern when properly built and ad- 455 www.americanradiohistory.com Figure 45 REAR VIEW OF TUNER SHOWING PLACEMENT OF MAJOR COMPON- ENTS Rotary inductor is drivby Johnson 116.2084 counter dial. Coaxial receptacle JI Input Is mounted directly below rotary inductor. n termination. The transmitter is turned on (preferably at reduced input) and resonance is established in the amplifier tank circuit. The sensitivity control of the tuner is adjusted to provide near full scale deflection on the bridge meter. Various settings of Si, L2, and Cl should be tried to obtain a reduction of bridge reading. As tuner resonance is approached, the meter reading will decrease and the sensitivity control should be advanced. When the system is in resonance, the meter will read zero. All loading adjustments may then be made with the transmitter controls. The tuner should be readjusted whenever the frequency of the transmitter is varied by an appreciable amount. ohm Figure 46 CLOSE -UP OF SWR BRIDGE Simple SWR bridge is mounted below the chassis of the tuner. Carbon resistors are mounted to two copper rings to form low inductance resistor. one -ohm Bridge capacitors form triangular configuration for lowest lead inductance. Balancing capacitor C2 is at lower right. www.americanradiohistory.com HANDBOOK Single -wire Antenna Tuner 453 52 a INPUT FROM XMTR R1 n 5 250 25 C2 1Q 5 SINGLE TUNE C1 330 2RV. SENSITIVITY Figure 43 ANTENNA TUNER IS HOUSED IN METAL CABINET 7' x 8" IN SIZE. TAP AT 15 T., 27 T., FROM POINT A INDUCTOR (10 NH) Inductance switch SI and sensitivity control are at left with (clockwise) position. The bridge is balanced when the input impedance of the tuner is 52 ohms resistive. This is the condition for maximum energy transfer between transmission line and antenna. The meter is graduated in arbitrary units, since actual SWR value is not required. Tuner Construction parts placement in tuner is shown in figures 43 and 45. Tapped coil L1 is mounted upon 1-inch ceramic insulators, and all major components are mounted above deck with the exception of the SWR bridge (figure 46). The components of the bridge are placed below deck, adjacent to the coaxial input plug mounted on the rear apron of the chassis. The ten 10 -ohm resistors are soldered to two 1 -inch rings made of copper wire as shown in the photograph. The bridge capacitors are attached to this assembly with extremely short leads.The 1N56 crystal mounts at right angles to the resistors to insure minimum amount of capacitive coupling between the resistors and the detector. The output lead from the bridge passes through a ceramic feed thru insulator to the top side of the chassis. Connection to the antenna is made by means of a large feedthru insulator mounted on the back of the tuner cabinet. This insulator is not visible in the photographs. Major the VARIABL E CI- JOHNSON 350E20 C2- CENTRALA8 J1 -TYPE SO -239 TYPE 822 RECEPTACLE R1-TEN 10-OHM -WATT CAR1 BON RESISTORS IN PARA- LLEL. INC TYPE LTA Figure 44 SCHEMATIC, SINGLE -WIRE ANTENNA TUNER counter dial for L2 at center. Output tuning capacitor CI is at right. SWR meter is mounted above SI. mum 'll = -1 L1- 35 TURNS e 18, 2- DIA., 3.9- LONG (A /R -DL/.e) L2- JOHNSON 229 -207 ANT 010V MICA 0 WIRE L2 Si Bridge The SWR bridge must be calibrated for 52 ohm service. This can be done by temporarily disconnecting the lead between the bridge and the antenna tuner and connecting a 2 -watt, 52 ohm carbon resistor to the junction of R1 and the negative terminal of the 1N56 diode. The opposite lead of the carbon resistor is grounded to the chassis of the bridge. A small amount of r-f energy is fed to the input of the bridge until a reading is obtained on the r -f voltmeter. The 25 mmfd bridge balancing capacitor C2 (see figure 46) is then adjusted with a fibre -blade screwdriver until a zero reading is obtained on the meter. The sensitivity control is advanced as the meter null grows, in order to obtain the exact point of bridge balance. When this point is found, the carbon resistor should be removed and the bridge attached to the antenna tuner. The bridge capacitor is sealed with a drop of nail polish to prevent misadjustment. Calibration All tuning adjustments are made to obtain proper transmitter loading with a balanced (zero meter reading) bridge condition. The tuner is connected to the transmitter through a random length of 52 ohm coaxial line, and the single wire antenna is attached to the output terminal of the tuner. Transmitter loading controls are set to approximate a 52 Tuner Adjustments www.americanradiohistory.com THE Antennas and Antenna Matching 452 PARALLEL -WIRE TO 40-e0 M. ANTENNA TO RECEIVER SINGLE WIRE ANTENNA COAX. LINES TO IOM. ANT 20M ANT r RADIO 55 IL 1 COAXIAL SWR INDICATOR LINE FROM XMTR .001 CERAMICS Figure 42 L ANTENNA TUNER AND SWR INDICATOR FOR RANDOM LENGTH HERTZ ANTENNA TO TRANSMITTER THROUGH HARMONIC FILTER Figure 41 ALTERNATIVE COAXIAL ANTENNA COUPLER This circuit is recommended not only as being most desirable when coaxial lines with low s.w.r. are being used to feed antenna systems such as rotatable beams, but when It also Is desired to feed through open -wire line to some sort of multi -band antenna for the lower frequency ranges. The tuned circuit of the antenna coupler is operative only when using the open -wire feed, and then It is In operation both for transmit and receive. in such an application will be found to be adequate, since harmonic attenuation has been accomplished ahead of the antenna coupler. However, the circuit will be easier to tune, although it will not have as great a bandwidth, if the operating Q is made higher. An alternative arrangement shown in figure 41 utilizes the antenna coupling tank circuit only when feeding the coaxial output of the transmitter to the open -wire feed line (or similar multi -band antenna) of the 40- 80 meter antenna. The coaxial lines to the 10 -meter beam and to the 20 -meter beam would be fed directly from the output of the coaxial antenna changeover relay through switch S. rise to several thousand ohms (near half -wave resonance) and the reactive component of the load can rapidly change from positive to negative values, or vice -versa. It is possible to match a 52 -ohm transmission line to such an antenna at almost any frequency between 1.8 me and 30 me with the use of a simple tuner of the type shown in figure 42. A variable series inductor L, and a variable shunt capacitor Cl permit circuit resonance and impedance transformation to be established for most antenna lengths. Switch S1 permits the selection of series capacitor C for those instances when the single wire antenna exhibits large values of positive reactance. To provide indication for the tuning of the network, a radio frequency bridge (SWR meter) is included to indicate the degree of mismatch (standing wave ratio) existing at the input to the tuner. All adjustments to the tuner are made with the purpose of reaching unity standing wave ratio on the coaxial feed system between the tuner and the transmitter. A simple antenna tuner for use with transmitters of 250 watts power or less is shown in figures 43 through 46. A SWR bridge circuit is used to indicate tuner resonance. The resistive arm of the bridge consists of ten 10 -ohm, 1 -watt carbon resistors connected in parallel to form a 1 -ohm resistor (R1). The other pair of bridge arms are capacitive rather than resistive. The bridge A Practical Antenna Tuner 22 -12 A Single -Wire Antenna Tuner One of the simplest and least expensive antennas for transmission and reception is the single wire, end -fed Hertz antenna. When used over a wide range of frequencies, this type of antenna exhibits a very great range of input impedance. At the low frequency end of the spectrum such an antenna may present a resistive load of less than one ohm to the transmitter, combined with a large positive or negative value of reactance. As the frequency of operation is raised, the resistive load may detector is a simple r -f voltmeter employing a 1N56 crystal diode and a 0 -1 d.c. milliammeter. A sensitivity control is incorporated to prevent overloading the meter when power is first applied to the tuner. Final adjustments are made with the sensitivity control at its maxi- www.americanradiohistory.com HANDBOOK Antenna ter, assuming that the antenna feed line is being operated with a low standing -wave ratio. However, there are many cases where it is desirable to feed a multi -band antenna from the output of the harmonic filter, where a tuned line is being used to feed the antenna, or where a long wire without a separate feed line is to be fed from the output of the harmonic filter. In such cases an antenna coupler is required. Some harmonic attenuation will be provided by the antenna coupler, particularly if it is well shielded. In certain cases when a pi network is being used at the output of the transmitter, the addition of a shielded antenna coupler will provide sufficient harmonic attenuation. But in all normal cases it will be necessary to include a harmonic filter between the output of the transmitter and the antenna coupler. When an adequate harmonic filter is being used, it will not be necessary in normal cases to shield the antenna coupler, except from the standpoint of safety or convenience. Function of an Antenna Coupler The function of the antenna coupler is, basically, to transform the impedance of the antenna system being used to the correct value of resistive impedance for the harmonic filter, and hence for the transmitter. Thus the antenna coupler may be used to resonate the feeders or the radiating portion of the antenna system, in addition to its function of imped- ance transformation. It is important to remember that there is nothing that can be done at the antenna coupler which will eliminate standing waves on the antenna transmission line. Standing waves are the result of reflection from the antenna, and the coupler can do nothing about this condition. However, the antenna coupler can resonate the feed line (by introducing a conjugate impedance) in addition to providing an impedance transformation. Thus, a resistive impedance of the correct value can be presented to the harmonic filter, as in figure 36, regardless of any reasonable value of standing -wave ratio on the antenna transmission line. All usual types of antenna couplers fall into two classifications: (1) inductively coupled resonant systems as exemplified by those shown in figure 39, and (2) conductively coupled pi- network systems such as shown in figure 40. The inductively -coupled system is much more commonly used, since it is convenient for feeding a balanced line from the coTypes of Antenna Couplers axial output of the usual harmonic filter. The pi- network system is most useful for feeding a length of wire from the output of a transmitter. Couplers COAX TO RECEIVER TRANSMITTER HARMONIC FILTER 451 SINGLE WIRE ANTENNA COAX ANT. CHANGEOVER RELAY Figure 40 PI- NETWORK ANTENNA COUPLER An arrangement such as illustrated above is convenient for feeding an end -fed Hertz antenna, or a random length of wire for portable or emergency operation, from the nominal value of impedance of the harmonic filter. Several general methods for using the inductively- coupled resonant type of antenna coupler are illustrated in figure 39. The coupling between the link coil L and the main tuned circuit need not be variable; in fact it is preferable that the correct link size and placement be determined for the tank coil which will be used for each band, and then that the link be made a portion of the plug -in coil. Capacitor C then can be adjusted to a pre- determined value for each band such that it will resonate with the link coil for that band. The reactance of the link coil (and hence the reactance of the capacitor setting which will resonate the coil) should be about 3 or 4 times the impedance of the transmission line between the antenna coupler and the harmonic filter, so that the link coupling circuit will have an operating Q of 3 or 4. The use of capacitor C to resonate with the inductance of the link coil L will make it easier to provide a low standing -wave ratio to the output of the harmonic filter, simply by adjustment of the antenna- coupler tank circuit to resonance. If this capacitor is not included, the system still will operate satisfactorily, but the tank circuit will have to be detuned slightly from resonance so as to cancel the inductive reactance of the coupling link and thus provide a resistive load to the output of the harmonic filter. Variations in the loading of the final amplifier should be made by the coupling adjustment at the final amplifier, not at the antenna coupler. The pi- network type of antenna coupler, as shown in figure 40 is useful for certain applications, but is primarily useful in feeding a single -wire antenna from a low- impedance transmission line. In such an application the operating Q of the pi network may be somewhat lower than that of a pi network in the plate circuit of the final amplifier of a transmitter, as shown in figure 38. An operating Q of 3 or 4 www.americanradiohistory.com 450 THE Antennas and Antenna Matching RADIO COAX. TO RECEIVER TRANSMITTER U HARMONIC FILTER LJ ¡ COAX ANT. CNANGEOVEq RELAY PARALLEL-WIRE LINE TO ANTENNA L SINGLE -WIRE HERTZ ANTENNA OR FEEDER CEPP FEEDERS O SINGLE-WIRE COAX. LINE ANTENNA OR FEEDLINE TO ANTENNA O © Figure 39 ALTERNATIVE ANTENNA -COUPLER CIRCUITS Plug -in coils, one or two variable capacitors of the split-stator variety, and a system of switches or plugs and jacks may be used in the antenna coupler to accomplish the feeding of different types of antennas and antenna transmission lines from the coaxial input line from the transmitter or from the antenna changeover relay. Link L should be resonated with capacitor C at the operating frequency of the transmitter so that the harmonic filter will operate into o resistive load impedance of the correct nominal value. ended output stage down to the 50 -ohm impedance of the usual harmonic filter and its sub - sequent load. In a pi network of this type the harmonic attenuation of the section will be adequate when the correct value of C, and L are being used and when the r e s on a n t dip in C, is sharp. If the dip in C, is broad, or if the plate current persists in being too high with C2 at maximum setting, it means that a greater value of capacitance is required at C2, assuming that the values of C, and L are correct. 22 -11 Antenna Couplers As stated in the previous section, an antenna coupler is not required when the impedance of the antenna transmission line is the same as the nominal impedance of the harmonic fil- www.americanradiohistory.com HANDBOOK Pi- Network Coupling Systems 449 COAX. TO RECEIVER i SHIELO HARMONIC oAK---- ATTENUATING C 1 COAX 1 ANT. CHAANGEOVER 1 - - -I 1 ANTENNA FEEDLINE OR CO ANTENNA COUPLER TO Figure 37 TUNED -LINK OUTPUT CIRCUIT Capacitor C should be adjusted so os to tune out the inductive reactance of the coupling link, L. Loading of the amplifier then is varied by physically varying the coupling between the plate tank of the final amplifier and the antenna coupling link, The pi- network coupling system offers two advantages: (1) a mechanical coupling variation is not required to vary the loading of the final amplifier, and (2) the pi network (if used with an operating Q of about 15) offers within itself a harmonic attenuation of 40 db or more, in addition to the harmonic attenuation provided by the additional harmonic attenuating filter. Some Pi- Network Coupling commercial equipments (such as the Collins amateur transmitters) incorporate an L network in addition to the pi network, for accomplishing the impedance transformation in two steps and to provide additional harmonic attenuation. Tuning the Pi- Section Coupler Tuning of a pi- network coupling circuit such as illustrated in figure 38 is accomplished in the following manner: First remove the connection between the output of the amplifier and the harmonic filter (load). Tune C2 to a capacitance which is large for the band in use, adding suitable additional ca- pacitance by switch S if operation is to be on one of the lower frequency bands. Apply reduced plate voltage to the stage and dip to resonance with C,. It may be necessary to vary the inductance in coil L, but in any event resonance should be reached with a setting of C, which is approximately correct for the desired value of operating Q of the pi network. Next, couple the load to t h e amplifier (through the harmonic filter), apply reduced plate voltage again and dip to resonance with C,. If the plate current dip with load is too low (taking into consideration the reduced plate voltage), decrease the capacitance of C2 and again dip to resonance, repeating the procedure until the correct value of plate current is obtained with full plate voltage on the stage. There should be a relatively small change required in the setting of C, (from the original setting of C, without load) if the operating Q of the network is correct and if a large value of impedance transformation is being employed-as would be the case when transforming from the plate impedance of a singleCOAX TO RECEIVER HARMONIC _ATTENUATING FILTER = r COAXIAL ANTENNA ,CHANGEOVER RELAY TO FCCDLINE -MOR ANTENNA COUPLER Figure 38 PI- NETWORK ANTENNA COUPLER The design of pi-network output circuits is discussed end shunting capacitors selected by switch S are for in Chapter Thirteen. The additional output use on the lower frequency ranges. Inductor L may be selected by a tap switch, it may be continuously variable, or plug -in inductors may be used. www.americanradiohistory.com 448 THE Antennas and Antenna Matching AT TRANSMITTER I EXCITER PORTION rI NAL AMPLIFIE COUPLING RDJUSTMENT HARMONIC ATTENUATING SYSTEM ANTENNA COUPLER RADIO AT ANTENNA IMPEDANCE TRANSMISSION LIN[ MATCHING TANTENNA RADIATING SYSTEM Figure 36 ANTENNA COUPLING SYSTEM The antenna coupling system illustrated above is for use when the antenna transmission line does not have the same characteristic impedance as the TVI filter, and when the standing -wave ratio on the antenna transmission line may or may not be low. within or adjacent to the antenna. The feed line coming down from the an t e n n a system should have a characteristic impedance equal to the nominal impedance of the harmonic filter, and the impedance matching at the antenna should be such that the standing -wave ratio on the antenna feed line is less than 2 to 1 over the range of frequency to be fed to the antenna. Such an arrangement may be used with open -wire line, ribbon or tubular line, or with coaxial cable. The use of coaxial cable is to be recommended, but in any event the impedance of the antenna transmission line should be the same as the nominal impedance of the harmonic filter. The arrangement of figure 35 is more or less standard for commercially manufactured equipment for amateur and commercial use in the h -f and v -h -f range. The arrangement of figure 36 merely adds an antenna coupler between the output of the harmonic attenuating filter and the antenna transmission line. The antenna coupler will have some harmonic- attenuating action, but its main function is to transform the impedance at the station end of the antenna transmission line to the nominal value of the harmonic filter. Hence the arrangement of figure 36 is more general than the figure 35 system, since the inclusion of the antenna coupler allows the system to feed an antenna transmission line of any reasonable impedance value, and also without regard to the standing -wave ratio which might exist on the antenna transmission line. Antenna couplers are discussed in a following section. It will be noticed by reference to both figure 35 and figure 36 that a box labeled Coupling Adjustment is included in the block Output Coupling Adjustment diagram. Such an element is necessary in the complete system to afford an adjustment in the value of load impedance presented to the tubes in the final amplifier stage of the transmitter. The impedance at the input terminal of the harmonic filter is established by the antenna, through its matching system and the antenna coupler, if used. In any event the impedance at the input terminal of the harmonic filter should be very close to the nominal impedance of the filter. Then the Coupling Adjustment provides means for transforming this impedance value to the correct operating value of load impedance which should be presented to the final amplifier stage. There are two common ways for accomplishing the antenna coupling adjustment, as illustrated in figures 37 and 38. Figure 37 shows the variable -link arrangement most commonly used in home -constructed equipment, while the pi- netowrk coup ling arrangement commonly used in commercial equipment is illustrated in figure 38. Either method may be used, and each has its advantages. Variable-Link Coupling variable -link method illustrated in figure 37 has the advantage that standard manufactured components m a y be used with no changes. However, for greatest bandwidth of operation of the coupling circuit, the reactance of the link coil, L, and the reactance of the link tuning capacitor, C, should both be beThe tween 3 and 4 times the nominal load impedance of the harmonic filter. This is to say that the inductive reactance of the coupling link L should be tuned out or resonated by capacitor C, and the operating Q of the L -C link circuit should be between 3 and 4. If the link coil is not variable with respect to the tank coil of the final amplifier, capacitor C may be used as a loading control; however, this system is not recommended since its use will require adjustment of C whenever a frequency change is made at the transmitter. If L and C are made resonant at the center of a band, with a link circuit Q of 3 to 4, and coupling adjustment is made by physical adjustment of L with respect to the final amplifier tank coil, it usually will be possible to operate over an entire amateur band without change in the coupling system. Capacitor C normally may have a low voltage rating, even with a high power transmitter, due to the low Q and low impedance of the coupling circuit. www.americanradiohistory.com Antenna Coupling Systems HANDBOOK When insulators are subject to very high r -f voltages, they should be cleaned occasionally if in the vicinity of sea water or smoke. Salt scum and soot are not readily dislodged by rain, and when the coating becomes heavy enough, the efficiency of the insulators is greatly impaired. If a very pretentious installation is to be made, it is wise to check up on both underwriter's rules and local ordinances which might be applicable. If you live anywhere near an airport, and are contemplating a tall pole, it is best to investigate possible regulations and ordinances pertaining to towers in the district, before starting construction. Coupling to the Antenna System 22 -10 When coupling an antenna feed system to a transmitter the most important considerations are as follows: (1) means should be provided for varying the load on the amplifier; (2) the two tubes in a push -pull amplifier should be equally loaded; (3) the load presented to the final amplifier should be resistive (non -reactive) in character; and (4) means should be provided to reduce harmonic coupling between the final amplifier plate tank circuit and the antenna or antenna transmission line to an extremely low value. The problem of coupling the power output of a high -frequency or v -h -f transmitter to the radiating portion of the antenna system has been materially complicated by the virtual necessity for eliminating interference to TV reception. However, the TVI- elimination portion of the problem may always be accomplished by adequate shielding of the transmitter, by filtering of the control and power leads which enter the transmitter enclosure, and by the inclusion of a harmonic -attenuating filter beThe TransmitterLoading Problem 1NIELD rE%CITER PORTION FINAL AMPLIFIER tween the output of the transmitter and the antenna system. Although TVI may be eliminated through inclusion of a filter between the output of a shielded transmitter and the antenna system, the fact that such a filter must be included in the link between transmitter and antenna makes it necessary that the transmitter -loading problem be re- evaluated in terms of the necessity for inclusion of such a filter. Harmonic- attenuating filters must be operated at an impedance level which is close to their design value; therefore they must operate into a resistive termination substantially equal to the characteristic impedance of the filter. If such filters are operated into an impedance which is not resistive and approximately equal to their characteristic impedance: (1) the capacitors used in the filter sections will be subjected to high peak voltages and may be damaged, (2) the harmonic- attenuating properties of the filter will be decreased, and (3) the impedance at the input end of the filter will be different from that seen by the filter at the load end (except in the case of the half-wave type of filter). It is therefore important that the filter be included in the transmitter-to-an tenna circuit at a point where the impedance is close to the nominal value of the filter, and at a point where this impedance is likely to remain fairly constant with variations in frequency. There are two basic arrangements which include all the provisions required in the transmitter-to- antenna coupling system, and which permit the harmonic -attenuating filter to be placed at a position in the coupling system where it can be operated at an impedance level close to its nominal value. These arrangements are illustrated in block -diagram form in figures 35 and 36. The arrangement of figure 35 is recommended for use with a single -band antenna system, such as a dipole or a rotatable array, wherein an impedance matching system is included Block Diagrams of Transmitter -to- Antenna Coupling Systems AT TRANSMITTER COUPLING Ap1U5TMENT _ HARMONIC TTENUATI SYSTEM 447 AT ANTENNA I TRANSMISSION LINE MPE DANCED RADIATING MATCHING T ANTENNAn SYSTEM Figure 35 ANTENNA COUPLING SYSTEM The harmonic suppressing antenna coupling system illustrated above is for use when the antenna transmission line has a low standing -wave ratio, and when the characteristic impedance of the antenna transmission line is the same as the nominal impedance of the low -pass harmonicattenuating filter. www.americanradiohistory.com 4 4 6 Antennas and Antenna M waiting for it to show excessive wear or deterioration. It is an excellent idea to tie both ends of the halyard line together in the manner of a flag -pole line. Then the antenna is tied onto the place where the two ends of the halyard are joined. This procedure of making the halyard into a loop prevents losing the top end of the halyard should the antenna break near the end, and it also prevents losing the halyard completely should the end of the halyard carelessly be allowed to go free and be pulled through the pulley at the top of the mast by the antenna load. A somewhat longer piece of line is required but the insurance is well worth the cost of the additional length of rope. Often a tall tree can be called upon to support one end of an antenna, but one should not attempt to attach anything to the top, as the swaying of the top of the tree during a heavy wind will complicate matters. If a tree is utilized for support, provision should be made for keeping the antenna taut without submitting it to the possibility of being severed during a heavy wind. This can be done by the simple expedient of using a pulley and halyard, with weights attached to the lower end of the halyard to keep the antenna taut. Only enough weight to avoid excessive sag in the antenna should be tied to the halyard, as the continual swaying of the tree submits the pulley and halyard to considerable wear. Galvanized iron pipe, or steel -tube conduit, is often used as a vertical radiator, and is quite satisfactory for the purpose. However, when used for supporting antennas, it should be remembered that the grounded supporting poles will distort the field pattern of a vertically polarized antenna unless spaced some distance from the radiating portion. Trees as Supports The life of a wood mast or pole can be increased several hundred per cent by protecting it from the elements with a coat or two of paint. And, of course, the appearance is greatly enhanced. The wood should first be given a primer coat of flat white outside house paint, which can be thinned down a bit to advantage with second -grade linseed oil. For the second coat, which should not be applied until the first is thoroughly dry, aluminum paint is not only the best from a preservative standpoint, but looks very well. This type of paint, when purchased in quantities, is considerably cheaper than might be gathered from the price asked for quarter-pint cans. Portions of posts or poles below the surface of the soil can be protected from termites and moisture by painting with creosote. While not Painting THE atching RADIO so strong initially, redwood will deteriorate much more slowly when buried than will the white woods, such as pine. The Antenna Wire antenna or array itself presents no especial problem. A few considerations should be borne in mind, however. For instance, soft -drawn copper should not be used, as even a short span will stretch several per cent after whipping around in the wind a few weeks, thus affecting the resonant frequency. Enameled -copper wire, as ordinarily available at radio stores, is usally soft drawn, but by tying one end to some object such as a telephone pole and the other to the frame of an auto, a few husky tugs can be given and the wire, after stretching a bit, is equivalent to hard drawn. Where a long span of wire is required, or where heavy insulators in the center of the span result in considerable tension, copper clad steel wire is somewhat better than harddrawn copper. It is a bit more expensive, though the cost is far from prohibitive. The use of such wire, in conjunction with strain insulators, is advisable, where the antenna would endanger persons or property should it break. For transmission l i n e s and tuning stubs steel -core or hard -drawn wire will prove awkward to handle, and soft -drawn copper should, therefore, be used. If the line is Ion g, the strain can be eased by supporting it at several points. More important from an electrical standpoint than the actual size of wire used is the soldering of joints, especially at current loops in an antenna of low radiation resistance. In fact, it is good practice to solder all joints, thus insuring quiet operation when the antenna is used for receiving. question that often arises is that of insulation. It depends, of course, upon the r-f voltage at the point at which the insulator is placed. The r -f voltage, in turn, depends upon the distance from a current node, and the radiation resistance of the antenna. Radiators having low radiation resistance have very high voltage at the voltage loops; consequently, better than usual insulation is advisable at those points. Open -wire lines operated as non -resonant lines have little voltage across them; hence the most inexpensive ceramic types are sufficiently good electrically. With tuned lines, the Insulation A voltage depends upon the amplitude of the standing waves. If they are very great, the voltage will reach high values at the voltage loops, and the best spacers available are none too good. At the current loops the voltage is quite low, and almost anything will suffice. www.americanradiohistory.com Antenna HANDBOOK Construction 445 2X4 Figure 34 SIMPLE WOOD MASTS Shown at (A) is the method of assembly, and at (B) is the completed structure, of the conventional "Aframe" antenna most. At (C) is shown a structure which is heavier but more stable than the A -frame for heights above about 40 feet. TWO SAWHORSES /-\\ Iv-S 2X2 CROSSPIECES GROUND LEVEL i I CONCRETE éd '9JJéi s' © if a gin pole about 20 feet high is installed about 30 or 40 feet to the rear of the direction in which the antenna is to be raised. A line from a pulley on the top of the gin pole is then run to the top of the pole to be raised. The gin pole comes into play when the center of the mast has been raised 10 to 20 feet above the ground and an additional elevated pull is required to keep the top of the mast coming up as the center is raised further above ground. Steel tubing masts of the telescoping variety are widely available at a moderate price for use in supporting television antenna arrays. These masts usually consist of several 10-foot lengths of electrical metal tubing (EMT) of sizes such that the sections will telescope. The 30 -foot and 40-foot lengths are well suited as masts for supporting antennas and arrays of the types used on the amateur bands. The masts are constructed in such a manner that the bottom 10-foot length may be guyed permanently before the other sections are raised. Then the upper sections may be extended, beginning with the top -mast section, until the mast is at full length(provided a strong wind is not blowing) following which all the guys may be anchored. It is important that there be no load on the top of the mast when the "vertical" raising method is to be employed. Using TV Masts Guy Wires Guy wires should never be pulled taut; a small amount of slack is desirable. Galvanized wire, somewhat heavier than seems sufficient for the job, should be used. The heavier wire is a little harder to handle, but costs only a little more and takes longer to rust through. Care should be taken to make sure that no kinks exist when the pole or tower is ready for erection, as the wire will be greatly weakened at such points if a kink is pulled tight, even if it is later straightened. If "dead men" are used for the guy wire terminations, the wire or rod reaching from the dead men to the surface should be of non -rusting material, such as brass, or given a heavy coating of asphalt or other protective substance to prevent destructive action by the damp soil. Galvanized iron wire will last only a short time when buried in moist soil. Only strain -type (compression) insulators should be used for guy wires. Regular ones might be sufficiently strong for the job, but it is not worth taking chances, and egg -type strain halyard insulators are no more ex- pensive. Only a brass or bronze p u l l e y should be used for the halyard, as a high pole with a rusted pulley is truly a sad affair. The bearing of the pulley should be given a few drops of heavy machine oil before the pole or tower is raised. The halyard itself should be of good material, preferably water-proofed. Hemp rope of good quality is better than window sash cord from several standpoints, and is less expensive. Soaking it thoroughly in engine oil of medium viscosity, and then wiping it off with a rag, will not only extend its life but minimize shrinkage in wet weather. Because of the difficulty of replacing a broken halyard it is a good idea to replace it periodically, without www.americanradiohistory.com 444 ance of 123 ohms. Z, is one -quarter wavelength long at the mid-frequency and has an impedance of 224 ohms. is the balanced line to be matched(in this case 300 ohms) and may be any length. 4 Other system parameters for different output and input impedances may be calculated from the following: Transformation ratio (r) for each section is: r= Z N Zrn Where N is the number of sections. In the above case, Z5 3 Z r Impedance between sections, as times the preceding section. Z=_, = r = r X Z,_, Z,_ X is r Z and Mid -frequency (m): m= F, + F, For 40 -20 -10 meters = +30 7 - 18.5 Mc. 2 and one -quarter wavelength = 12 feet. For 20 -10-6 meters -- 14 + 54 - 34 Mc. 2 and one -quarter wavelength = 5.5 feet. The impedances of the sections are: Z, =V Z, Z, = Z4 -7- Zo= X V Z:-, Z,_, X RADIO been concerned primarily with the electrical characteristics and considerations of antennas. Some of the physical aspects and mechanical problems incident to the actual erection of antennas and arrays will be discussed in the following section. Up to 60 feet, there is little point in using mast -type antenna supports unless guy wires either must be eliminated or kept to a minimum. while a little more difficult to erect, because of their floppy nature, fabricated wood poles of the type to be described will be just as satisfactory as more rigid types, provided many guy wires are used. Rather expensive when purchased through the regular channels, 40- and 50 -foot telephone poles sometimes can be obtained quite reasonably. In the latter case, they are hard to beat, inasmuch as they require no guying if set in the ground six feet (standard depth), and the resultant pull in any lateral direction is not in excess of a hundred pounds or so. For heights of 80 to 100 feet, either three sided or four -sided lattice type masts are most practicable. They can be made self- supporting, but a few guys will enable one to use a smaller cross section without danger from high winds. The torque exerted on the base of a high self- supporting mast is terrific dur- ing a strong wind. 2 Z3-4 \7Z74.727 XZ, Generally, the larger number of taper sections the greater will be the bandwidth of the system. 22 -9 THE Antennas and Antenna Matching Antenna Construction The foregoing portion of this chapter has The "A- Frame" Figures 34A and 34B show the standard method of conMost struction of the A -frame type of mast. This type of mast is quite frequently used since there is only a moderate amount of work involved in the construction of the assembly and since the material cost is relatively small. The three pieces of selected 2 by 2 are first set up on three sawhorses or boxes and the holes drilled for the three 1/4inch bolts through the center of the assembly. Then the base legs are spread out to about 6 feet and the bottom braces installed. Then the upper b r aces and the cross pieces are installed and the assembly given several coats of good -quality paint as a protection against weathering. Figure 34C shows another common type of mast which is made up of sections of 2 by 4 placed end -to -end with stiffening sections of 1 by 6 bolted to the edge of the 2 by 4 sections. Both types of masts will require a set of top guys and another set of guys about one third of the way down from the top. Two guys spaced about 90 to 100 degrees and pulling against the load of the antenna will normally be adequate for the top guys. Three guys are usually used at the lower level, with one directly behind the load of the antenna and two more spaced 120 degrees from the rear guy. The raising of the mast is made much easier www.americanradiohistory.com Matching Systems HANDBOOK Ì` - L FEET = _ Center to Center Spacing in Inches 234 Zo= FIMCI Impedance in Ohms for %z" Diameters for I/4 " Diameters 170 188 207 225 248 1.25 1.5 1.75 Z2 Impedance in Ohms 1 Q MATCHING SECTION TUBING L 46B F Mc 443 2 250 277 298 318 335 PARALLEL TUBING SURGE IMPEDANCE FOR So MATCHING SECTIONS z The Collins Transmission Line UNTUNED LINE ANY LENGTH ally can be obtained by either designing or adjusting the matching section for a dipole to a fo r transmitters a r e numerous; however this output system becomes awkward when it is desired to feed an antenna system utilizing a balanced i n p u t. For some time the Collins Radio Co. has been experimenting with a balun and tapered line system for matching a coaxial output transmitter to an open -wire balanced transmission line. Considerable success has been obtained and matching systems good over a frequency range as great as four to one have been developed. Illustrated in figure 33 is one type of matching system which is proving satisfactory over this range. Z, is the transmitter end of the system and may be any length of 52 -ohm coaxial cable. Z2 is one quarter wavelength long at the mid-frequency of the range to be covered and is made of 75 ohm coaxial cable. ZA is a quarter- wavelength shorted section of cable at the mid -frequency. ZD (ZA and Z2) forms a 200-ohm quarter -wave section. The ZA section is formed of a conductor of the same diameter as Z2. The difference in length between ZA and Z2 is accounted for by the fact that Z2 is a coaxial conductor with a solid dielectric, whereas the dielectric for ZD is air. Z2 is one -quarter wavelength long at the mid -frequency and has an impedMatching System Figure 32 HALF -WAVE RADIATOR FED BY "Q BARS" The Q matching section is simply o quarter wave transformer whose impedance is equal to the geometric mean between the impedance at the center of the antenna and the impedance of the transmission line to be used to feed the bottom of the transformer. The transformer may be made up of parallel tubing, a four -wire line, or any other type of transmission line which has the correct value of impedance. have The advantage of unbalanced output networks surge impedance that is the geometric mean between the line impedance and 72 ohms, the latter being the theoretical radiation re- sistance of a half -wave doublet either infinitely high or a half wave above a perfect ground. Though the radiation resistance may depart somewhat from 72 ohms under actual condi- tions, satisfactory results will be obtained with this assumed value,so long as the dipole radiator is more than a quarter wave above effective earth, and reasonably in the clear. The small degree of standing waves introduced by a slight mismatch will not increase the line losses appreciably, and any small amount of reactance present can be tuned out at the transmitter termination with no bad effects. If the reactance is objectionable, it may be minimized by making the untuned line an integral number of quarter waves long. A Q- matched system can be adjusted precisely, if desired, by constructing a matching section to the calculated dimensions with provision for varying the spacing of the Q section conductors slightly, after the untuned line has been checked for standing waves. Zo ZJ Z4 If__,_,,./____n____ Zz ZI ---- 2J=123R v ZA ZS I Z= 22411 Z5=30011 (ANY LENGTH) } INNER L OUTER CONDUCTORS SHORTED AT EACH END Figure 33 COLLINS TRANSMISSION LINE MATCHING SYSTEM wide-band system for matching a 52 -ohm coaxial line to a balanced 300 -ohm line over a 4:7 wide frequency range. A www.americanradiohistory.com 442 The open stub should be resonated in the same manner as the shorted stub before attaching the transmission line; however, in this case, it is necessary to prune the stub to resonance, as there is no shorting bar. Sometimes it is handy to have a stub hang from the radiator to a point that can be reached from the ground, in order to facilitate adjustment of the position of the transmission -line attachment. For this reason, a quarter -wave stub is sometimes made three -quarters wavelength long at the higher frequencies, in order to bring the bottom nearer the ground. Operation with any odd number of quarter waves is the same as for a quarter -wave stub. Any number of half waves can be added to either a quarter-wave stub or a half -wave stub without disturbing the operation, though losses and frequency sensitivity will be lowest if the shortest usable stub is employed. See figure 31. Stub Length (Electrical) %.etc. 1/4- % -1 t wavelengths Y2 -1 -1 i5.2 -etc . wavelength s Linear THE Antennas and Antenna Matching R -F Transformers Current -Fed Radiator Voltage -Fed Radiator Open Stub Shorted Stub Shorted Stub Open Stub resonant quarter -wave line the unusual property of acting much as a transformer. Let us take, for example, a section consisting of no. 12 wire spaced 6 inches, which happens to have a surge impedance of 600 ohms. Let the far end be terminated with a pure resistance, and let the near end be fed with radio frequency energy at the frequency for which the line is a quarter wavelength long. If an impedance measuring set is used to measure the impedance at the near end while the impedance at the far end is varied, an interesting relationship between the 600 -ohm characteristic surge impedance of this particular quarter -wave matching line, and the impedance at the ends will be discovered. When the impedance at the far end of the line is the same as the characteristic surge impedance of the line itself (600 ohms), the impedance measured at the near end of the quarter -wave line will also be found to be A has 600 ohms. Under these conditions, the line would not have any standing waves on it, since it is terminated in its characteristic impedance. Now, let the resistance at the far end of the line be doubled, or changed to 1200 ohms. The impedance measured at the near end of the line will be found to have been cut in RADIO half, to 300 ohms. If the resistance at the far end is made half the original value of 600 ohms or 300 ohms, the impedance at the near end doubles the original value of 600 ohms, and becomes 1200 ohms. As one resistance goes up, the other goes down proportionately. It will always be found that the characteristic surge impedance of the quarter -wave matching line is the geometric mean between the impedance at both ends. This relationship is shown by the following formula: ZMS = ZA ZL where = Impedance of matching = Antenna resistance. = Line impedance. ZMS ZA ZL section. The impedance inverting char- Quarter -Wave Matching Transformers acteristic of a quarter -wave section of transmission line is widely used by making such a section of line act as a quarter-wave transformer. The Johnson Q feed system is a widely known application of the quarter-wave transformer to the feeding of a dipole antenna and array consisting of two dipoles. However, the quarter -wave transformer may be used in a wide number of applications wherever a transformer is required to match two impedances whose geometric mean is somewhere between perhaps 25 and 750 ohms when transmission line sections can be used. Paralleled coaxial lines may be used to obtain the lowest impedance mentioned, and open -wire lines composed of small conductors spaced a moderate distance may be used to obtain the higher impedance. A short list of impedances, which may be matched by quarter -wave sections of transmission line having specified impedances, is given below. Load or Ant. Impedance 1 20 30 50 75 100 300 77 95 110 150 173 480 600 98 120 139 190 220 110 134 155 212 r Feed -Line Impedance Quarter Wave Transformer Impedance 245 The standard form of Johnson n feed to a doublet is shown in figure 32. An impedance match is obtained by utilizing a matching section, the surge impedance of which is the geometric mean between the transmission line surge impedance and the radiation resistance of the radiator. A sufficiently good match usuJohnson -Q Feed System www.americanradiohistory.com HANDBOOK Matching Systems 441 section. If they are not used, the T- section will detune the dipole when the T- section is attached to it. The two capacitors may be ganged together, and once adjusted for minimum detuning action, they may be locked. A suitable housing should be devised to protect these capacitors from the weather. Additional information on the adjustment of the T -match is given in the chapter covering rotary beam antennas. SHORTING BAR ANTENNA unbalanced version of the T -match may be used to feed a dipole from an unbalanced coaxial line. Such a device is called a Gamma Match, and is illustrated in figure 30. The length of the Gamma rod and the spacing of it from the dipole determine the impedance level at the transmission line end of the rod. The series capacitor is used to tune out the reactance introduced into the system by the Gamma rod. The adjustment of the Gamma Match is discussed in the chapter covering rotary beam antennas. The "Gamma" Match An connecting a resonant section of transmission line (called a matching stub) to either a voltage or current loop and attaching parallel -wire non resonant feeders to the resonant stub at a suitable voltage (impedance) point, standing waves on the line may be virtually eliminated. The stub is made to serve as an auto- transformer. Stubs are particularly adapted to matching an open line to certain directional arrays, as will be described later. Matching Stubs SHORTING BAR NON-RESONANT FEEDERS ANTENNA FEEDER TAPS NEAR END OF STUB NON- RESONANT FEEDERS STUB When the When a stub is used to current feed a radiator, the stub should either be left open at the bottom end instead of shorted, or else made a half wave long. Current Feed STUB By stub attaches to the antenna at a voltage loop, the stub should be a quarter wavelength long electrically, and be shorted at the bottom end. The stub can be resonated by sliding the shorting bar up and down before the non -resonant feeders are attached to the stub, the antenna being shock -excited from a separate radiator during the process. Slight errors in the length of the radiator can be compensated for by adjustment of the stub if both sides of the stub are connected to the radiator in a symmetrical manner. Where only one side of the stub connects to the radiating system, as in the Zepp and in certain antenna arrays, the radiator length must be exactly right in order to prevent excessive unbalance in the untuned line. A dial lamp may be placed in the center of the shorting stub to act as an r -f indicator. Voltage Feed 11 OPEN t7 STUB NON- RESONANTSHORTING BAR FEEDER Figure 31 MATCHING -STUB APPLICATIONS An end -fed half-wave antenna with a quarterwave shorted stub is shown at (A). (B) shows the use of a half-wave shorted stub to feed a relatively low impedance point such as the center of the driven element of a parasitic array, or the center of a half-wave dipole. The use of an open -ended quarter -wave stub to feed a low impedance is illustrated at (C). (D) shows the conventional use of a shorted quarter -wave stub to voltage feed two half-wave antennas with a 180. phase difference. www.americanradiohistory.com 440 THE Antennas and Antenna Matching RADIO DRIVEN ELEMENT MOVEABLE CLAMP GAMMA ROD RESONATING CONDENSER -.:7°-LE.D- I7 EL.T-T0. 0. 50 -70 OHM COAXIAL FEED LINE I TD irrN-LE.D IE1 l000HU. ANY LENGTN Figure 30 SOIVEN ELEMENT GAMMA MATCH FOR CONNECTING AN UNBALANCED COAXIAL LINE TO A THE BALANCED DRIVEN ELEMENT / / SOD-.00 OHM ECLDC03 l00-.00 ON Figure 29 FOLDED -ELEMENT MATCHING SYSTEMS Drawing (A) above shows a half-wave made up to two parallel wires. If one of the wires is broken as in (B) and the feeder connected, the feed -point impedance is multiplied by four; such an antenna is commonly called a "folded doublet." The feed-point impedance for a simple half -wave doublet fed in this manner is approximately 300 ohms, depending upon antenna height. Drawing (C) shows how the feed -point impedance can be multiplied by a factor greater than four by making the half of the element that is broken smaller in diameter than the unbroken half. An extension of the principles of (B) and (C) is the arrangement shown at (D) where the section into which the feeders are connected is considerably shorter than the driven element. This system is most convenient when the driven element is too long (such as for o 28 -Mc. or 14 -Mc. array) for a convenient mechanical arrangement of the system shown at (C). wire of such a radiator, as shown in figure 29, the effective feed-point resistance of the antenna or array will be increased by a factor of N2 where N is equal to the number of conductors, all in parallel, of the same diameter in the array. Thus if there are two conductors of the same diameter in the driven element or the antenna the feed -point resistance will be multiplied by 22 or 4. If the antenna has a radiation resistance of 75 ohms its feed -point resistance will be 300 ohms, this is the case of the conventional folded- dipole as shown in figure 29B. If three wires are used in the driven radiator the feed -point resistance is increased by a factor of 9; if four wires are used the impedance is increased by a factor of 16, and so on. In certain cases when feeding a parasitic array it is desirable to have an impedance step up different from the value of 4:1 obtained with two elements of the same diameter and 9:1 with three elements of the same diameter. Intermediate values of impedance step up may be obtained by using two elements of different diameter for the complete driven element as shown in figure 29C. If the conductor that is broken for the feeder is of smaller diameter than the other conductor of the radiator, the impedance step up will be greater than 4:1. On the other hand if the larger of the two elements is broken for the feeder the impedance step up will be less than 4:1. "T" method of matching a balanced low- impedance transmission line to the driven element of a parasitic array is the T match illustrated in figure 29D. This method is an adaptation of the multi -wire doublet principle which is more practicable for lower-frequency parasitic axrays such as those for use on the 14 -Mc. and 28 -Mc. bands. In the system a section of tubing of approximately one-half the diameter of the driven element is spaced about four inches below the driven element by means of clamps which hold the T- section mechanically and which make electrical connection to the driven element. The length of the T- section is normally between 15 and 30 inches each side of the center of the dipole for transmission lines of 300 to 600 ohms impedance, assuming 28 -Mc. operation. In series with each leg of the T- section and the transmission line is a series resonating capacitor. These two capacitors tune out the reactance of the TThe Match A Matching Systems HANDBOOK -I D MATCHING SECTION 439 transmitter end of the feed line which will change the magnitude of the standing waves on the antenna transmission line. The delta type matched -impedance antenna system is Delta- Marched Antenna System shown in figure 28. The im- NON-RESONANT LINE Figure 28 THE DELTA- MATCHED DIPOLE ANTENNA The dimensions for the portions of the an. tenno ore given in the text. between three types of transmission line: (1) Ribbon or tubular molded 300 -ohm line is widely used up to moderate power levels (the "transmitting" type is useable up to the kilowatt level). (2) Open -wire 400 to 600 ohm line is most commonly used when the antenna is some distance from the transmitter, because of the low attenuation of this type of line. (3) Coaxial line (usually RG -8 /U with a 52 -ohm characteristic impedance) is widely used in v -h -f work and also on the lower frequencies where the feed line must run underground or through the walls of a building. Coaxial line also is of assistance in TVI reduction since the r -f field is entirely enclosed within the line. Molded 75 -ohm line is sometimes used to feed a doublet antenna, but the doublet has been largely superseded by the folded -dipole antenna fed by 300 -ohm ribbon or tubular line when an antenna for a single band is required. discussed earlier, standing waves on the antenna transmission line, in the transmitting case, are a result of reflection from the point where the feed line joins the antenna system. The magnitude of the standing waves is deterStanding Waves As was mined by the degree of mismatch between the characteristic impedance of the transmission line and the input impedance of the antenna system. When the feed -point impedance of the antenna is resistive and of the same value as the characteristic impedance of the feed line, standing waves will not exist on the feeder. It may be well to repeat at this time that there is no adjustment which can be made at the pedance of the transmission line is transformed gradually into a higher value by the fanned -out Y portion of the feeders, and the Y portion is tapped on the antenna at points where the antenna impedance is a compromise between the impedance at the ends of the Y and the impedance of the unfanned portion of the line. The constants of the system are rather critical, and the antenna must resonate at the operating frequency in order to minimize standing waves on the line. Some slight readjustment of the taps on the antenna is desirable, if appreciable standing waves persist in appearing on the line. The constants for a doublet are determined by the following formulas: L feet - 467.4 F megacycles 175 Dfeet F megacycles Efeet 147.6 F megacycles Where L is antenna length; D is the distance in from each end at which the Y taps on; E is the height of the Y section. Since these constants are correct only for a 600 -ohm transmission line, the spacing S of the line must be approximately 75 times the diameter of the wire used in the transmission line. For no. 14 B & S wire, the spacing will be slightly less than 5 inches. This system should never be used on either its even or odd harmonics, as entirely different constants are required when more than a single half wavelength appears on the radiating portion of the system. Multi -Wire Doublets When a doublet antenna or the driven element in consists of more than one wire or tubing conductor the radiation resistance of the antenna or array is increased slightly as a result of the increase in the effective diameter of the element. Further, if we split just one an array www.americanradiohistory.com 438 THE Antennas and Antenna Matching RADIO VOLTAGE CURVES 40 MET R5 20vMETERS IO METERS \% 8 AN TOS MSC -TOP 87 FT. VIA 33.5 ro !CwLtILMIw wo wwL*oRa FT. APE, Z' HIGH CENTER ANTENNA WIRE FEEDER Figure 27 SINGLE- WIRE -FED ANTENNA FOR ALL BAND OPERATION An antenna of this type for 40 -, 20- and 10meter operation would have a radiator 67 feet long, with the feeder tapped 11 feet off center. The feeder can be 33, 66 or 99 feet long. The some type of antenna for 80 -, 40 -, 20- and 10 -meter operation would have a radiator 134 feet long, with the feeder topped 22 feet off center. The feeder can be either 66 or 132 feet long. This system should be used only with those coupling methods which provide good harmonic suppression. JOA« ALL a2 Om. cODUL wiAEST SASE L.wE L«[ Figure 26 MECHANICAL CONSTRUCTION METER DISCONE OF 20- pending upon the wire size and the point of attachment to the antenna. The earth losses are comparatively low over ground of good conductivity. Since the single wire feeder radiates, it is necessary to bring it away from the antenna at right angle s to the antenna wire for at least one -half the length of the antenna. The correct point for best impedance match on the fundamental frequency is not suitable for harmonic operation of the antenna. In addition, the correct length of the antenna for fundamental operation is not correct for harmonic operation. Consequently, a compromise must be made in antenna length and point of feeder connection to enable the single -wirefed antenna to operate on more than one band. Such a compromise introduces additional reactance into the single wire feeder, and might cause loading difficulties with pi- network transmitters. To minimize this trouble, the single wire feeder should be made a multiple of 33 feet long. Two typical single -wire -fed antenna systems are shown in figure 27 with dimensions for multi -band operation. 22 -8 Matching Non -Resonant Lines to the Antenna Present practice in regard to the use of transmission lines for feeding antenna systems on the amateur bands is about equally divided www.americanradiohistory.com HANDBOOK Frequency Low i F 3.3 S 437 .x.o... ...........11 ..1.... M.^¡E S`4.0 D Discone .,I............ 30 II O i I I II l.s CC I I m I I H I z 25 I I e Io 14 le 22 26 30 34 3e 42 4e 50 54 5e FREQUENCY (Mc) II II I Figure 25 CURVE FOR A 13.2 MC. DISCONE ANTENNA. SWR IS BELOW 1.5 TO FROM II SWR 1 13.0 MC. TO 58 MC. R 32 OHM COAXIAL FEED LINE DIMENSIONS 20,15.11,10,4 METERS D. 12 L. S=10- R. is, 1e H.15' 7. 15, 1.10,6 METERS 1 =e S. e. D L= 12' R =12, H =10.5 11,10.6,2 METERS =e D= L=Ye- S R=ee H=e'3^ Figure 24 DIMENSIONS OF LOW- FREQUENCY DISCONE ANTENNA FOR LOW FREQUENCY CUTOFF AT 13.2 MC., 20.1 MC., AND 26 MC. The Discone is a vertically polarized radiator, producing an omnidirectional pattern similar to a ground plane. Operation on several amateur bands with low SWR on the coaxial feed line is possible. Additional information on L -F Discone by W2RYI in July, 1950 CQ magazine. of the radials may be reduced to 25 feet. As with all multi -band antennas that employ no lumped tuned circuits, this antenna offers no attenuation to harmonics of the transmitter. When operating on the lower frequency band, it would be wise to check the transmitter for second harmonic emission, since this antenna will effectively radiate this harmonic. The Low - Frequency Discone it discone antenna is widely used on the v -h -f bands, but until recently The has not been put to any great use on the lower frequency bands. Since the discone is a broad -band device, it may be used on several harmonically related amateur bands. Size is the limiting factor in the use of a discone, and the 20 meter band is about the lowest practical frequency for a discone of reasonable dimensions. A discone designed for 20 meter operation may be used on 20, 15, 11, 10 and 6 meters with excellent results. It affords a good match to a 50 ohm coaxial feed system on all of these bands. A practical discone antenna is shown in figure 24, with a SWR curve for its operation over the frequency range of 13 -55 Mc. shown in figure 25. The discone antenna radiates a vertically polarized wave and has a very low angle of radiation. For v -h -f work the discone is constructed of sheet metal, but for low frequency work it may be made of copper wire and aluminum angle stock. A suitable mechanical layout for a low frequency di scone is shown in figure 26. Smaller versions of this antenna may be constructed for 15, 11, 10 and 6 meters, or for 11, 10, 6 and 2 meters as shown in the chart of figure 24. For minimum wind resistance, the top "hat" of the discone is constructed from three -quarter inch aluminum angle stock, the rods being bolted to an aluminum plate at the center of the structure. The tips of the rods are all connected together by lengths of no. 12 enamelled copper wire. The cone elements are made of no. 12 copper wire and act as guy wires for the discone structure. A very rigid arrangement may be made from this design; one that will give no trouble in high winds. A 4" x 4" post can be used to support the discone structure. The discone antenna may be fed by a length of 50 -ohm coaxial cable directly from the transmitter, with a very low SWR on all bands. The old favorite single -wirefed antenna system is quite satisfactory for an impromptu all band antenna system. It is widely used for portable installations and "Field Day" contests where a simple, multi -band antenna is required. A single wire feeder has a characteristic impedance of some 500 ohms, deThe Single -WireFed Antenna www.americanradiohistory.com 436 THE Antennas and Antenna Matching of transmission line of any characteristic impedance into a feeder system such as this and the impedanc' at the far end of the line will be exactly the same value of impedance which the half -was e line sees at its termination. Hence this has been done in the antenna system shown in figure 22; an electrical half wave of line has been inserted between the feed point of the antenna and the 300 -ohm transmission line to the transmitter. The characteristic impedance of this additional half -wave section of transmission line has been made about 715 ohms (no. 20 wire spaced 6 inches), but since it is an electrical half wave long at 7 Mc. and operates into a load of 300 ohms at the antenna the 300 -ohm Twin -Lead at the bottom of the half-wave section still sees an impedance of 300 ohms. The additional half -wave section of transmission line introduces a negligible amount of loss since the current flowing in the section of line is the same which would flow in a 300 -ohm line at each end of the half -wave section, and at all other points it is less than the current which would flow in a 300 -ohm line since the effective impedance is greater than 300 ohms in the center of the half -wave section. This means that the loss is less than it would be in an equivalent length of 300 -ohm TwinLead since this type of manufactured transmission line is made up of conductors which are equivalent to no. 20 wire. So we see that the added section of 715 -ohm line has substantially no effect on the operation of the antenna system on the 7 -Mc. band. However, when the flat top of the antenna is operated on the 3.5-Mc. band the feed-point impedance of the flat top is approximately 3500 ohms. Since the section of 715 -ohm transmission line is an electrical quarter-wave in length on the 3.5-Mc. band, this section of line will have the effect of transforming the approximately 3500 ohms feed-point impedance of the antenna down to an impedance of about 150 ohms which will result in a 2:1 standing -wave ratio on the 300 -ohm Twin -Lead transmission line from the transmitter to the antenna system. The antenna system of figure 22 operates with very low standing waves over the entire 7 -Mc. band, and it will operate with moderate standing waves from 3500 to 3800 kc. in the 3.5-Mc. band and with sufficiently low standing -wave ratio so that it is quite usable over the entire 3.5 -Mc. band. This antenna system, as well as all other types of multi -band antenna systems, must be used in conjunction with some type of harmonic- reducing antenna tuning network even though the system does present a convenient impedance value on both bands. RADIO L 300 OHM OPEN -WIRE TV TYPE LINE I6Á -e0 METERS TO' L. V=52' 60 -40 METERS L =33' V =2s' /*P-6 RADIALS 5 OHM COAX IA INE Figure 23 THE MULTEE TWO -BAND ANTENNA This compact antenna can be used with excellent results on 160/80 and 80/40 meters. The feedline should be held as vertical as possible, since it radiates when the antenna is operated on its fundamental frequency. The "Multee" Antenna An antenna that works well on 160 and 80 meters, or 80 and 40 meters and is sufficiently compact to permit erection on the average city lot is the W68CX Multee antenna, illustrated in figure 23. The antenna evolves from a vertical two wire radiator, fed on one leg only. On the low frequency band the top portion does little radiating, so it is folded down to form a radiator for the higher frequency band. On the lower frequency band, the antenna acts as a top loaded vertical radiator, while on the higher frequency band, the flattop does the radiating rather than the vertical portion. The vertical portion acts as a quarter wave linear transformer, matching the 6000 ohm antenna impedance to the 50 ohm impedance of the coaxial transmission line. The earth below a vertical radiator must be of good conductivity not only to provide a low resistance ground connection, but also to provide a good reflecting surface for the waves radiated downward towards the ground. For best results, a radial system should be installed beneath the antenna. For 160 -80 meter operation, six radials 50 feet in length, made of no. 16 copper wire should be buried just below the surface of the ground. While an ordinary water pipe ground system with no radials may be used, a system of radials will provide a worthwhile increase in signal strength. For 80 -40 meter operation, the length www.americanradiohistory.com HANDBOOK Multi -band Antennas 435 144 33' OR NV LONG- 400 01.163 OPEN -WIRE TV TYPE LINE ANTENNA TUNER OR MATCH NOX SMAL LINE Figure 21 MULTI - BAND ANTENNA USING FAN DIPOLE TO LIMIT IMPEDANCE EXCURSIONS ON HARMONIC FREQUENCIES - wire spaced 4 to 6 inches the antenna system is sometimes called a center -fed zepp. With this type of feeder the impedance at the transmitter end of the feeder varies from about 70 ohms to approximately 5000 ohms, the same as is encountered in an end -fed zepp antenna. This great impedance ratio requires provision for either series or parallel tuning of the feeders at the transmitter, and involves quite high r -f voltages at various points along the feed 14 line. If the feed line between the transmitter and the antenna is made to have a characteristic impedance of approximately 300 ohms the excursions in end -of- feeder impedance are greatly reduced. In fact the impedance then varies from approximately 75 ohms to 1200 ohms. With this much lowered impedance variation it is usually possible to use series tuning on all bands, or merely to couple the antenna directly to the output tank circuit or the harmonic reduction circuit without any separate feeder tuning provision. There are several practicable types of transmission line which can give an impedance of approximately 300 ohms. The first is, obviously, 300 -ohm Twin -Lead. Twin -Lead of the receiving type may be used as a resonant feed line in this case, but its use is not recommended with power levels greater than perhaps 150 watts, and it should not be used when lowest loss in the transmission line is desired. For power levels up to 250 watts or so, the transmitting type tubular 300 -ohm line may be used, or the open -wire 300 -ohm TV line may be employed. For power levels higher than this, a 4- wire transmission line, or a line built of one -quarter inch tubing should be used. FOLDED -TOP Figure 22 DUAL -BAND ANTENNA Even when a 300 -ohm transmission line is used, the end-of- feeder impedance may reach a high value, particularly on the second harmonic of the antenna. To limit the impedance excursions,, a two -wire flat -top may be employed for the radiator, as shown in figure 21. The use of such a radiator will limit the impedance excursions on the harmonic frequencies of the antenna and make the operation of the antenna matching unit much less critical. The use of a two -wire radiator is highly recommended for any center -fed multi -band antenna. Folded Flot -Top Dual -Band Antenna As has been mentioned earlier, there is an increasing tendency among amateur operators to utilize rotary or fixed arrays for the 14-Mc. band and those higher in frequency. In order to afford complete coverage of the amateur bands it is then desirable to have an additional system which will operate with equal effectiveness on the 3.5 -Mc. and 7 -Mc. bands, but this low- frequency antenna system will not be required to operate on any bands higher in frequency than the 7 -Mc. band. The antenna system shown in figure 22 has been developed to fill this need. This system consists essentially of an open -line folded dipole for the 7 -Mc. band with a special feed system which allows the antenna to be fed with minimum standing waves on the feed line on both the 7 -Mc. and 3.5 -Mc. bands. The feed -point impedance of a folded dipole on its fundamental frequency is approximately 300 ohms. Hence the 300 -ohm Twin Lead shown in figure 22 can be connected directly into the center of the system for operation only on the 7 -Mc. band and standing waves on the feeder will be very small. However, it is possible to insert an electrical half -wave www.americanradiohistory.com 4 34 RADIO THE Antennas and Antenna Matching -lao - r L.90' FOR /0 -40 IMETES 1551014 LIN[ If fOOR VIED FOR 1-2 /MI IMPEDANCE AT TNt TRANSMITTER END OF TOE iP BANDS LI La TOPE OF TuNINO LiNt if AP02. L2 1200 ONMf 50 orHc OPERATION PARALLEL 11. SS PALACELI 250 LIMP n5 Figure A 19 TWO -BAND MARCONI ANTENNA 160 -80 METER OPERATION 105 FOR 7 !di RS T Since this antenna type is an unbalanced radiating system, its use is not recommended with high -power transmitters where interference to broadcast listeners is likely to be encountered. The r-f voltages encountered at the end of zepp feeders and at points an electrical half wave from the end are likely to be quite high. Hence the feeders should be supported an adequate distance from surrounding objects and sufficiently in the clear so that a chance encounter between a passerby and the feeder is unlikely. The coupling coil at the transmitter end of the feeder system should be link coupled to the output of the low -pass TVI filter in order to reduce harmonic radiation. The Two -Band Marconi Antenna A three- eighths wavelength antenna may be operated on its harmonic Marconi frequency, providing good two band performance from a simple wire. Such an arrangement for operation on 160 -80 meters, and 80 -40 meters is shown in figure 19. On the fundamental (lowest) frequency, the antenna acts as a three- eighths wavelength series -tuned Marconi. On the second harmonic, the antenna is a current -fed three -quarter wavelength antenna operating against ground. For proper operation, the antenna should be resonated on its second harmonic by means of a grid-dip oscillator to the operating frequency most used on this particular band. The Q of the antenna is relatively low, and the antenna will perform well over a frequency range of several hundred kilocycles. The overall length of the antenna may be varied slightly to place its self- resonant frequency in the desired region. Bends or turns in the antenna tend to make it resonate higher in frequency, and it may be necessary to lengthen it a bit to resonate it at the chosen frequency. For fundamental operation, the series condenser is inserted in the circuit, and the antenna may be resonated to any point in the lower frequency band. As with any Marconi n MRCS PA PARALLEL 5ERIEs Mc MC stain PARALLEL SI S] PA 14 MC 211 MC PARALLEL SI 1100 Ow. 1100 ON25 M 0.045 w Tf 00uf taw 0005 0021 A 75 OHMS /200 OHMS 55 1200 1200 o02í 1200 ONMs Mc Mc 1200 ONMf 1100 OHMS 100 1200 OHMS Haw Ms Ms :00 CENTER-FED ANTENNA DIMENSIONS Figure 20 CENTER -FED MULTI BAND ANTENNA FOR - type antenna, the use of a good ground is essential. This antenna works well with transmitters employing coaxial antenna feed, since its transmitting impedance on both bands is in the neighborhood of 40 to 60 ohms. It may be attached directly to the output terminal of such transmitters as the Collins 32V and the Viking H. The use of a low -pass TVI filter is of course recommended. For multi -band operation, the center fed antenna is without doubt the best compromise. It is a balanced system on all bands, it requires no ground return, and when properly tuned has good rejection properties for the higher harmonics generated in the transmitter. It is well suited for use with the various multi -band 150 -watt transmitters that are currently so popular. For proper operation with The Center -Fed Multi -Band Antenna these transmitters, an antenna tuning unit must be used with the center-fed antenna. In fact, some sort of tuning unit is necessary for any type of efficient, multi -band antenna. The use of such questionable antennas as the "offcenter fed15 doublet is an invitation to TVI troubles and improper operation of the transmitter. A properly balanced antenna is the best solution to multi -band operation. When used in conjunction with an antenna tuning unit, it will perform with top efficiency on all of the major amateur bands. Several types of center -fed antenna systems are shown in figure 20. If the feed line is made up in the conventional manner of no. 12 or no. HANDBOOK `[ rant+ NH... Multi -band Antennas a 6' L I11 L N' L 411' 100 OMM O 433 FEEDER SPREADERS 'OR 3310 NC AND 7114 NC rpm 7I0O NC AND 1310 NC. rpA I4100 NC. AND a= MC N* '"ITEM 600 A LINE THE Figure 15 THREE -QUARTER WAVE DOUBLET This antenna arrangement FOLDED SHORTED END 600 OHM LINE TO TRANSMITTER L. will give very the fundamental frequency and with 1 . - 496 S FT effective radiator on the second harmonic but the pattern of radiation will be different from that on the fundamental, and the standing -wave ratio on the feed line will be greater. The flat top of the antenna must be made of open wire rather than ribbon or tubular line. For greater operating convenience, the shorting switch may be replaced with a section of transmission line. If this transmission line is made one -quarter wavelength long for the fundamental frequency, and the free end of the line is shorted, it will act as an open circuit across the center insulator. At the second harmonic, the transmission line is one -half wavelength long, and reflects the low impedance of the shorted end across the center insulator. Thus the switching action is automatic as the frequency of operation is changed. Such an installation is shown in figure 16. The end -fed Hertz antenna shown in figure 17 is not as effective a radiating system as Figure 16 AUTOMATIC BANDSWITCHING STUB FOR THE THREE -QUARTER WAVE FOLDED DOUBLET The antenna of Figure 15 may be used with a shorted stub line in place of the switch normally used for second harmonic operation. types, but it is particularit is desired to install an for a test, or for field -day of the radiator should be clear as possible. In any event at least three quarters of the total wire length should be in the clear. Dimensions for optimum operation on various amateur bands are given in addition in figure 17. many other antenna ly convenient when antenna in a hurry work. The flat top as high and in the The end- f ed Zepp has long been a favorite for multi -band operation. It is shown in figure 18 along with recommended dimensions for operation on various amateur band groups. The End -Fed Zepp - LI r01÷ .1ta 3.5 AND 7 MC. 3.9 MC. AND 26 MC. Ll36 a.1 MC SS MC L= 137' 3,5 11C L136 1 LO 20' 7 OR1 Sr1IAD[RS all MC MC SILO.- L1 TTn Dr 1n a $10,01 MRALL[L 137 n $1 14 BANDS 3.3, 7, 14 AND 26 MC. 3.5, 7 AND 14 MC. MC FROM xMTR. 17 END-FED ZEPP RECOMMENDED LENGTHS FOR THE END FED HERTZ www.americanradiohistory.com FIGURE 18 TIMING 110610/ M1ALL11L /ARALLIL M1 MIES LINK Figure FT the switch closed on twice frequency. The End -Fed Hertz 67 FT WHEN ANTENNA IS 195 FT. - 96 Fr. L' 33 FT L ' 6 satisfactory operation with a 600 -ohm feed line for operation with the switch open on - -- - - 432 THE Antennas and Antenna Matching 3 RADIO - ANTENNA eO METER7 L13sC 4001lUr PNENOUC !LOCKS SEE 3.5 3.1 3.7 30 3 rIC.12 40 SWR C200YLr Figure 13 CURVE OF 80 -METER BROAD-BAND INNUR CONDUCTOR NOT USED SEE FIG.12 FOR CONNECTION DIPOLE 52 OHM COAXIAL LINE Figure ohms. The ground losses are now reduced by a factor of 4. In addition, the antenna may be directly fed from a 50 -ohm coaxial line, or directly from the unbalanced output of a pi- network transmitter. Since a certain amount of power may still be lost in the ground connection, it is still of greatest importance that a good, low resistance ground be used with this antenna. Shown in figures 11 and 12 are broad -band dipoles for the 40 and 80 meter amateur bands, designed by Collins Radio Co. for use with the Collins 32V -3 and KW -1 transmitters. These fan -type dipoles have excellent broad-band response, and are designed to be fed with a 52 -ohm unbalanced coaxial line, making them suitable for use with many of the other modem transmitters, such as the Barker and Williamson 5100, Johnson Ranger, and Viking. The antenna system consists of a fan -type dipole, a balun matching section, and a suitable coaxial feedline. The Q of the half -wave 80 meter doublet is lowered by decreasing the effective length -todiameter ratio. The frequency range of operation of the doublet is increased considerably by this change. A typical SWR curve for the 80 meter doublet is shown in figure 13. The balanced doublet is matched to the unbalanced coaxial line by the one -quarter wave balun. If desired, a shortened balun may be used (figure 14). The short balun is capacity loaded at the junction between the balun and the broad-band dipole. The Collins Brood -bond Dipole System 22 -7 Multi -Band Antennas The availability of a multi -band antenna is great operating convenience to an amateur station. In most cases it will be found best to install an antenna which is optimum for the band which is used for the majority of the a M L7'3- 4.0 FREQUENCY (MC) 14 SHORT BALUN FOR 40 AND 80 METERS available operating time, and then to have an additional multi -band antenna which may be pressed into service for operation on another band when propagation conditions on the most frequently used band are not suitable. Most amateurs use, or plan to install, at least one directive array for one of the higher- frequency bands, but find that an additional antenna which may be used on the 3.5-Mc. and 7.0 -Mc. band, or even up through the 28 -Mc. band is almost indispensable. The choice of a multi -band antenna depends upon a number of factors such as the amount of space available, the band which is to be used for the majority of operating with the antenna, the radiation efficiency which is desired, and the type of antenna tuning network to be used at the transmitter. A number of recommended types are shown in the next pages. The 3í -Wave Folded Doublet Figure 15 type which be very shows an antenna will be found to effective when a moderate amount of space is available, when most of the operating will be done on one band with occasional operation on the second harmonic. The system is quite satisfactory for use with high -power transmitters since a 600 ohm non -resonant line is used from the antenna to the transmitter and since the antenna system is balanced with respect to ground. With operation on the fundamental frequency of the antenna where the flat top is % wave long the switch SW is left open. The system affords a very close match between the 600ohm line and the feed point of the antenna. Kraus has reported a standing -wave ratio of approximately 1.2 to 1 over the 14 -Mc. band when the antenna was located approximately one -half wave above ground. For operation on the second harmonic the switch SW is closed. The antenna is still an www.americanradiohistory.com HANDBOOK 1 110' 44.9 r 1ldllllllr FOR 11.10.10c FOR DETAIL SEE FIG. A PHENOLIC BLOCK 2XI.5XC WRAP CABLES AND BLOCK WITH SCOTCH ELECTRICALTAPE SPACE BLOCKS 0' APART ALONG BALUN +j// 2 X 1.3 X 0.5 ENOLIC BLOCK / // //%//// //// /// 431 411111110 THE TWO W IRES MAY BE SPREAD EITHER HORIZONTALLYOR VERTICALLY. WRAP CABLES AND BLOCK ITN SCOTCH ELECTRICAL T SPACE BLOCKS B'APART ALONG BALUN i iW.N.I '-11.11.IuIrI 114 OS- rI.Y 0.5W LS- FIGURE A FIGURE B FIGURE A CUT OFF SHIELD AND OUTER JACKET AS SHOWN. ALLOW DIELECTRIC TO E %TEND PART WAY TO OTHER CABLE. COVER ALL EXPOSED SHIELD AND DIELECTRIC ON BOTH CABLES WITH A CONTINUOUS WRAPPING OF SCOTCH ELECTRICAL TAPE TO EXCLUDE MOISTURE. B KEEP BALUN AT LEAST CLEAR OF GROUND AND OTHER OBJECTS. FOR DETAIL SEE FIG A - Antennas Conserving Space DETAIL SEE FIGURE B 52 OHM RG-8/U, ANY LENGTH REMOVE OUTER JACKET FROM A SHORT LENGTH OF CABLE AS SHOWN HERE. UNBRAID THE SHIELD OF COAX CUT OFF THE DI- ELECTRIC AND INNER CONDUCTOR FLUSH WITH THE OUTER JACKET. DO HOT CUT THE SHIELD. WRA SHIELD DIELECTRIC ON CONNECTION. BEING VERY CAREFUL NOT TO DAMAGE THE DIELECTRIC MATERIAL. HOLD CABLE O STRAIGHT WHILE SOLDERING. COVER THE AREA WITH A CONTINUOUS WRAPPING OF SCOTCH ELECTRICAL TAPE. NO CONNECTION TO INNER CONDUC- KEEP BALUN AT LEAST OF COAX C AROUND SHIELD OF COAX D. SOLDER THE FIGURE B REMOVE OUTER JACKET FROM A SHORT LENGTH OF CABLE AS SHOWN HERE. UNBRAID THE SHIELD OF COAX C CUTOFF THE DIELECTI(IC AND INNER CON DUCTOR FLUSH WITH THE OUTER JACKET. DO NOT CUT THE SHIELD. WRAP SHIELD OF COAX C AROUND SHIELD OF COAX D. SOLDER THE CONNECTION. BEING VERY CAREFUL NOT TO DAMAGE THE DIELECTRIC MATERIAL. HOLD CABLE D STRAIGHT WHILE SOLDERING. COVER CUT OFF SHIELD AND OUTER JACKET AS SHOWN. ALLOW DIELECTRIC TO EXTEND PART WAY TO OTHER CABLE. COVER ALL EXPOSED SHIELD AND BOTH CABLES WRAP- PING OF SCOTCH ELECTRICAL TAPE TO EXCLUDE MOISTURE. OF GROUND AND OTHER FOR B- LEAR OBJECTS. DETAIL SEE FIGURE B THE AREA WITHACONTIN- 52 OHM UOUS WRAPPING OF SCOTCH ELECTRICAL TAPE. N0 CONNEC T ION TO INNER CONDUCTORS. RD -B /U, ANY LENGTH TORS. DIMENSIONS SHOWN NERE ARE FOR THE 40 METER BAND. THIS ANTENNA MAY BE BUILT FOR OTHER BANDS BY US/Ni DIMENSIONS THAT ARE MULTIPLES OR SUBMUL TIPLES OF THE DIMENSIONS SHOWN. BALUN SPACING IS S. ON ALL BANDS. Figure 11 HALF -WAVE ANTENNA WITH QUARTER WAVE UNBALANCED TO BALANCED TRANSFORMER (BALUN) FEED SYSTEM FOR 40-METER OPERATION sions in terms of frequency are given on the drawing. An antenna of this type is 93 feet long for operation on 3600 kc. and 86 feet long for operation on 3900 kc. This type of antenna has the additional advantage that it may be operated on the 7 -Mc. and 14 -Mc. bands, when the flat top has been cut for the 3.5 -Mc. band, simply by changing the position of the shorting bar and the feeder line on the stub. A sacrifice which must be made when using a shortened radiating system, as for example the types shown in figure 9, is in the bandwidth of the radiating system. The frequency range which may be covered by a shortened antenna system is approximately in proportion to the amount of shortening which has been employed. For example, the antenna system shown in figure 9C may be operated over the range from 3800 kc. to 4000 kc. without serious standing waves on the feed line. If the b DIMENSIONS SHOWN HERE ARE FOR THE METER BAND. THIS ANTENNA MAY BE BUILT FOR OTHER BANDS BY USINE DIMENSIONS THAT ARE MULTIPLES OR SUBMUL TIPLES OF THE DIMENSIONS SHOWN. BALUN SPACING IS /.5. ON ALL BANDS. Figure 12 BROADBAND ANTENNA WITH QUARTER WAVE UNBALANCED TO BALANCED TRANSFORMER (BALUN) FEED SYSTEM FOR 80 -METER OPERATION antenna had been made full length it would be possible to cover about half again as much frequency range for the same amount of mismatch on the extremes of the frequency range. The Twin -Lead Marconi Antenna Much of the power loss in the Marconi antenna is a result of low radiation resistance and high ground resistance. In some cases, the ground resistance may even be be higher than the radiation resistance, causing a loss of 50 per cent or more of the transmitter power output. If the radiation resistance of the Marconi antenna is raised, the amount of power lost in the ground resistance is proportionately less. If a Marconi antenna is made out of 300 ohm TV -type ribbon line, as shown in figure 10, the radiation resistance of the antenna is raised from a low value of 10 or 15 ohms to a more reasonable value of 40 to 60 www.americanradiohistory.com 430 THE Antennas and Antenna Matching .. .,. AT Lo..,T RADIO /MM., 300 OHM -RIBBON- LINE YrLCGaa areeoaXa WIRES SHORTED TOGETHER AT END 52A COAXIAL TEED LINE Figure 10 TWIN -LEAD MARCONI ANTENNA FOR THE 80 AND 160 METER BANDS Figure 9 THREE EFFECTIVE SPACE CONSERVING ANTENNAS The arrangements shown at (A) and (B) are satisfactory where resonant feed line can be used. However, non- resonant 75 -ohm feed line may be used in the arrangement at (A) when the dimensions in wavelengths are as shown. In the arrangement shown at (B) low standing waves will be obtained on the feed line when the overall length of the antenna is a half wave. The arrangement shown at (C) may be tuned for any reasonable length of flat top to give a minimum of standing waves on the transmission line. quarter wavelength can be lengthened electrically by means of a series loading coil, and used as a quarter -wave Marconi. However, if the wire is made shorter than approximately one -eighth wavelength, the radiation resistance will be quite low. This is a special problem in mobile work below about 20 -Mc. 22 -6 Space -Conserving Antennas In many cases it is desired to undertake a considerable amount of operation on the 80meter or 40 -meter band, but sufficient space is simply not available for the installation of a half-wave radiator for the desired frequency of operation. This is a common experience of apartment dwellers. The shortened Marconi antenna operated against a good ground can be used under certain conditions, but the shortened Marconi is notorious for the production of broadcast interference, and a good ground connection is usually completely unobtainable in an apartment house. Essentially, the problem in producing an antenna for lower frequency operation in restricted space is to erect a short radiator which is balanced with respect to ground and which is therefore independent of ground for its operation. Several antenna types meeting this set of conditions are shown in figure 9. Figure 9A shows a conventional center -fed doublet with bent -down ends. This type of antenna can be fed with 75-ohm Twin -Lead in the center, or it may be fed with a resonant line for operation on several bands. The overall length of the radiating wire will be a few per cent greater than the normal length for such an antenna since the wire is bent at a position intermediate between a current loop and a voltage loop. The actual length will have to be determined by the cut-and -try process because of the increased effect of interfering objects on the effective electrical length of an antenna of this type. Figure 9B shows a method for using a two wire doublet on one half of its normal operating frequency. It is recommended that spaced open conductor be used both for the radiating portion of the folded dipole and for the feed line. The reason for this recommendation lies in the fact that the two wires of the flat top are not at the same potential throughout their length when the antenna is operated on one half frequency. Twin -Lead may be used for the feed line if operation on the frequency where the flat top is one -half wave in length is most common, and operation on one -half frequency is infrequent. However, if the antenna is to be used primarily on one -half frequency as shown, it should be fed by means of an open-wire line. If it is desired to feed the antenna with a non -resonant line, a quarter -wave stub may be connected to the antenna at the points X, X in figure 9B. The stub should be tuned and the transmission line connected to it in the normal manner. The antenna system shown in figure 9C may be used when not quite enough length is available for a full half-wave radiator. The dimen- www.americanradiohistory.com HANDBOOK Marconi Antenn a LOADING COILS Figure 429 `MAT a 7 8 LOADING THE MARCONI ANTENNA The various loading systems are discussed in the accom- panying text. O current flows through a r e s i s t o r, or if the ground itself presents some resistance, there will be a power loss in the form of heat. Improving the ground connection, therefore, provides a definite means of reducing this power loss, and thus increasing the radiated power. The best possible ground consists of as many wires as possible, each at least a quarter wave long, buried just below the surface of the earth, and extending out from a common point in the form of radials. Copper wire of any size larger than no. 16 is satisfactory, though the larger sizes will take longer to disintegrate. In fact, the radials need not even be buried; they may be supported just above the earth, and insulated from it. This arrangement is called a counterpoise, and operates by virtue of its high capacitance to ground. If the antenna is physically shorter than a quarter wavelength, the antenna current is higher, due to lower radiation resistance. Consequently, the power lost in resistive soil is greater. The importance of a good ground with short, inductive -loaded Marconi radiators is, therefore, quite obvious. With a good ground system, even very short (one- eighth wavelength) antennas can be expected to give a high percentage of the efficiency of a quarterwave antenna used with the same ground system. This is especially true when the short radiator is top loaded with a high Q (low loss) coil. Water-Pipe Grounds Water pipe, because of its corn - paratively large surface and cross section, has a relatively low r -f resistance. If it is possible to attach to a junction of several water pipes (where they branch in several directions and run for some distance under ground), a satisfactory ground connection will be obtained. If one of the pipes attaches to a lawn or garden sprinkler system in the immediate vicinity of the antenna, the effectiveness of the system will approach that of buried copper radials. The main objection to water-pipe grounds © © 0 0 0 is the possibility of high resistance joints in the pipe, due to the "dope" put on the coupling threads. By attaching the ground wire to a junction with three or more legs, the possibility of requiring the main portion of the r -f current to flow through a high resistance connection is greatly reduced. The presence of water in the pipe adds nothing to the conductivity; therefore it does not relieve the problem of high resistance joints. Bonding the joints is the best insurance, but this is, of course, impracticable where the pipe is buried. Bonding together with copper wire the various water faucets above the surface of the ground will improve the effectiveness of a water -pipe ground system hampered by high -resistance pipe couplings. antenna is an odd of electrical quarter waves long (usually only one quarter wave in length), and is always resonated to the operating frequency. The correct loading of the final amplifier is accomplished by varying the coupling, rather than by detuning the antenna from resonance. Physically, a quarter -wave Marconi may be made anywhere from one - eighth to three-eighths wavelength overall, meaning the total length of the antenna wire and ground lead from the end of the antenna to the point where the ground lead attaches to the junction of the radials or counterpoise wires, or where the water pipe enters the ground. The longer the antenna is made physically, the lower will be the current flowing in the ground connection, and the greater will be the overall radiation efficiency. However, when the antenna length exceeds three -eighths wavelength, the antenna becomes difficult to resonate by means of a series capacitor, and it begins to take shape as an end -fed Hertz, requiring a method of feed such as a pi network. A radiator physically much shorter than a Marconi Marconi A Dimensions number www.americanradiohistory.com 428 THE Antennas and Antenna Matching RADIO used for the radiator. Such an ant e nn a is shown in figure 6. The loaded ground-plane tends to have a rather high operating Q and operates only over a narrow band of frequencies. An operating range of about 100 kilo- cycles with a low SWR is possible on 80 meters. Operation over a larger frequency range is possible if a.higher standing wave ratio is tolerated on the transmission line. The radiation resistance of a loaded 80 -meter groundplane is about 15 ohms. A quarter wavelength (45 feet) of 52 -ohm coaxial line will act as an efficient feed line, presenting a load of approximately 180 ohms to the transmitter. 22 -5 COAX. PROM TRANS. Figure 7 FEEDING A QUARTER -WAVE MARCONI ANTENNA When an open -wire line is to be used, it may be link coupled to o series- resonant circuit between the bottom end of the Marconi and ground, as of (A). Alternatively, a reasonably good impedance match may be obtained between 52 -ohm coaxial line and the bottom of a resonant quarter -wave antenna, as illustrated at (B) above. The Marconi Antenna A grounded quarter-wave Marconi antenna, widely used on frequencies below 3 Mc., is sometimes used on the 3.5-Mc. band, and is also used in v -h -f mobile services where a compact antenna is required. The Marconi type antenna allows the use of half the length of wire that would be required for a half -wave Hertz radiator. The ground acts as a mirror, in effect, and takes the place of the additional quarter -wave of wire that would be required to reach resonance if the end of the wire were not returned to ground. The fundamental practical form of the Marconi antenna system is shown in figure 7. Other Marconi antennas differ from this type primarily in regard to the method of feeding the energy to the radiator. The feed method shown in figure 7B can often be used to advantage, particularly in mobile work. Variations on the basic Marconi antenna are shown in the illustrations of figure 8. Figures 8B and 8C show the "L" -type and "T "type Marconi antennas. These arrangements have been more or less superseded by the toploaded forms of the Marconi antenna shown in figures 8D, 8E, and 8F. In each of these latter three figures an antenna somewhat less than one quarter wave in length has been loaded to increase its effective length by the insertion of a loading coil at or near the top of the radiator. The arrangement shown at figure 8D gives the least loading but is the most practical mechanically. The system shown at figure 8E gives an intermediate amount of loading, while that shown at figure 8F, utilizing a "hat" just above the loading coil, gives the greatest amount of loading. The object of all the top -loading methods shown is to produce an increase in the effective length of the radiator, and thus to raise the point of maximum current in the radiator as far as pos- sible above ground. Raising the maximum -current point in the radiator above ground has two desirable results: The percentage of low angle radiation is increased and the amount of ground current at the base of the radiator is reduced, thus reducing the ground losses. To estimate whether a loading coil will probably be required, it is necessary only to note if the length of the antenna wire and ground lead is over a quarter wavelength; if so, no loading coil is needed, provided the series tuning capacitor has a high maximum capacitance. Amateurs primarily interested in the higher frequency bands, but who like to work 80 meters occasionally, can usually manage to resonate one of their antennas as a Marconi by working the whole system, feeders and all, against a water pipe ground, and resorting to a loading coil if necessary. A high- frequencyrotary, zepp, doublet, or single- wire -fed antenna will make quite a good 80 -meter Marconi if high and in the clear, with a rather Long feed line to act as a radiator on 80 meters. Where two-wire feeders are used, the feeders should be tied together for Marconi operation. Importance of Ground Connection With a quarter -wave antenna and a ground, the antenna current generally is measured with a meter placed in the antenna circuit close to the ground connection. If this www.americanradiohistory.com HANDBOOK Vertical Antennas 427 LOADING COIL APROXI MAYFLY D! TURNS RIZ WIRE, .S" DIAMETER AND RADIALS EACH la ground is an effective transmitting antenna for low-angle radiation, where ground conditions in the vicinity of the antenna are good. Such an antenna is not good for short -range sky wave communication, such as is the normal usage of the 3.5 -Mc. amateur band, but is excellent for short-range ground-wave communication such as on the standard broadcast band and on the amateur 1.8 -Mc. band. The vertical antenna normally will cause greater BO than an equivalent horizontal antenna, due to the much greater ground -wave field intensity. Also, the vertical antenna is poor for receiving under conditions where man -made interference is severe, since such interference is predominantly vertically polarized. Three ways of feeding a half -wave vertical antenna from an untuned transmission line are illustrated in figure 4. The J -fed system shown in figure 4A is obviously not practicable except on the higher frequencies where the extra length for the stub may easily be obtained. However, in the normal case the ground-plane vertical antenna is to be recommended over the J -fed system for high frequency work. 22 -4 An The Ground Plane Antenna effective low angle radiator for any ama- EACH 52 OHM COAXIAL LINE 45 FEET LONG VERTICAL WHIP Figure 5 THE LOW -FREQUENCY GROUND PLANE ANTENNA The radials o f the ground plane antenna should lie in a horizontal plane, although slight departures from this caused by nearby objects is allowable. The whip may be mounted on a short post, or on the roof of a building. The wire radials may slope downwards towards their tips, acting as guy wires for the installation. FOOT LONG RADIALS 52 OHM COAXIAL LINE, CENTER CONDUCTOR CONNECTS TO I Figure 6 80 METER LOADED GROUND PLANE ANTENNA Number of turns in loading coil to be adjusted until antenna system resonates at desired frequency in 80 meter band. teur band is the ground-plane antenna, shown in figure 5. So called because of the radial ground wires, the ground -plane antenna is not affected by soil conditions in its vicinity due to the creation of an artificial ground system by the radial wires. The base impedance of the ground plane is of the order of 30 to 35 ohms, and it may be fed with 52 -ohm coaxial line with only a slight impedance mis -match. For a more exact match, the ground-plane antenna may be fed with a 72 -ohm coaxial line and a quarter-wave matching section made of 52 -ohm coaxial line. The angle of radiation of the ground -plane antenna is quite low, and the antenna will be found less effective for contacts under 1000 miles or so on the 80 and 40 meter bands than a high angle radiator, such as a dipole. However, for DX contacts of 1000 miles or more, the ground -plane antenna will prove to be highly effective. The 80 -Meter Loaded Ground -Plane A vertical antenna of 66 feet in height presents quite a problern on a small lot, as the supporting guy wires will tend to take up quite a large portion of the lot. Under such conditions, it is possible to shorten the length of the vertical radiator of the ground plane by the inclusion of a loading coil in the vertical whip section. The ground-plane antenna may be artificially loaded in this manner so that a 25 -foot vertical whip may be www.americanradiohistory.com 426 THE Antennas and Antenna Matching RADIO 462 FMc tDZ FNC. I -- ' o.A.: oaPaii 1 FED VERTICAL 300 -OHM RIBBON STVB-FED VERTICAL © L-C FED VERTICAL 404 FNc 300 -ONM RIBBON 30 FMC. FOLDED Figure 3 DIPOLE WITH SHORTING STRAPS The impedance match and bandwidth char- acteristics ofa folded dipole maybe improved by shorting the two wires of the ribbon a distance out from the center equal to the velocity factor of the ribbon times the half -length of the dipole as shown at (A). An alternative arrangement with bent down ends for space conservation is illustrated at (13). times over the radiation resistance of the element, have both contributed to the frequent use of the multi -wire radiator as the driven element in a parasitic antenna array. Delta-Matched Doublet and Standard Doublet These two types of radiating elements are shown in figure 2L and figure 2M. The delta- matched doublet is described in detail in Section 22 -8 of this chapter. The standard doublet, shown in figure 2M, is fed in the center by means of 75ohm Twin -Lead, either the transmitting or the receiving type, or it may be fed by means of twisted -pair feeder or by means of parallel wire lamp -cord. Any of these types of feed line will give an approximate match to the center impedance of the dipole, but the 75ohm Twin -Lead is far to be preferred over the other types of low -impedance feeder due to the much lower losses of the polyethylene - dielectric transmission line. The coaxial- cable -fed doublet shown in figure 2N is a variation on the system shown in figure 2M. Either 52 -ohm coaxial cable or 75ohm coaxial cable may be used to feed the center of the dipole, although the 75 -ohm type Figure 4 HALF -WAVE VERTICAL ANTENNA SHOWING ALTERNATIVE METHODS OF FEED will give a somewhat better impedance match at normal antenna heights. Due to the asymmetry of the coaxial feed system difficulty may be encountered with waves traveling on the outside of the coaxial cable. For this reason the use of Twin -Lead is normally to be preferred over the use of coaxial cab e for feeding the center of a half-wave dipole. 1 Off- Center Fed Doublet shown in figure 2(0) is sometimes used to The system feed a half -wave dipole, especially when it is desired to use the same antenna on a number of harmonically-related frequencies. The feeder wire (no. 14 enamelled wire should be used) is tapped a distance of 14 per cent of the total length of the antenna either side of center. The feeder wire, operating against ground for the return current, has an impedance of approximately 600 ohms. The system works well over highly conducting ground, but will introduce rather high losses when the antenna is located above rocky or poorly conducting soil. The off -center fed antenna has a further disadvantage that it is highly responsive to harmonics fed to it from the transmitter. The effectiveness of the antenna system in radiating harmonics is of course an advantage when operation of the antenna on a number of frequency bands is desired. But it is necessary to use a harmonic filter to insure that only the desired frequency is fed from the transmitter to the antenna. 22 -3 The Half -Wave Vertical Antenna The half-wave vertical antenna with its bottom end from 0.1 to 0.2 wavelength a www.americanradiohistory.com bo v e THE Multi -wire Doublets RADIO series with the antenna coil or in parallel with it. A series tuning c a p a c i tor can be placed in series with one feeder leg without unbalancing the system. The tuned -doublet antenna is shown in figure 2D. The antenna is a current-fed system when the radiating wire is a half wave long electrically, or when the system is operated on its odd harmonics, but becomes a voltage fed radiator when operated on its even harin monics. The antenna has a different radiation pattern when operated on its harmonics, as would be expected. The arrangement used on the second harmonic is better known as the Franklin colinear array and is described in Chapter Twenty- three. The pattern is similar toa 1,j-wave dipole except that it is sharper in the broadside direction. On higher harmonics of operation there will be multiple lobes of radiation from the system. Figures 2E and 2F show alternative arrangements for using an untuned transmission line between the transmitter and the tuned-doublet radiator. In figure 2E a half -wave shorted line is used to resonate the radiating system, while in figure 2F a quarter-wave open line is utilized. The adjustment of quarter -wave and half -wave stubs is discussed in Section 19 -8. Doublets with Quarter -Wave Transformers The average value of feed impedance for a center -fed halfwave doublet is 75 ohms. The actual value varies with height and is shown in Chapter Twenty-one. Other methods of matching this rather low value of impedance to a medium -impedance transmission line are shown in (G), (H), and (I) of figure 2. Each of these three systems uses a quarter -wave transformer to accomplish the impedance transformation. The only difference between the three systems lies in the type of transmission line used in the quarter-wave transformer. (G) shows the Johnson Q system whereby a line made up of 1/2-inch dural tubing is used for the low- impedance linear transformer. A line made up in this manner is frequently called a set of Q bars. Illustration (H) shows the use of a four-wire line as the linear transformer, and (I) shows the use of a piece of 150 -ohm Twin -Lead electrically 1/2wave in length as the transformer between the center of the dipole and a piece of 300 -ohm Twin -Lead. In any case the impedance of the quarter -wave transformer will be of the order of 150 to 200 ohms. The use of sections of transmission line as linear transformers is discussed in detail in Section 22 -8. Multi -Wire Doublets alternative method for increasing the feed-point impedance of a dipole so that a medium -impedAn 425 ance transmission line may be used is shown in figures 2J and 2K. This system utilizes more than one wire in parallel for the radiating element, but only one of the wires is broken for attachment of the feeder. The most common arrangement uses two wires in the flat top of the antenna so that an impedance multiplication of four is obtained. The antenna shown in figure 2J is the socalled Twin -Lead folded dipole which is a commonly used antenna system on the mediumfrequency amateur bands. In this arrangement both the antenna and the transmission line to the transmitter are constructed of 300 -ohm Twin-Lead. The flat top of the antenna is made slightly less than the conventional length (462 /FMc, instead of 468 /FMc, for a single -wire flat top) and the two ends of the Twin -Lead are joined together at each end. The center of one of the conductors of the Twin -Lead flat top is broken and the two ends of the Twin -Lead feeder are spliced into the flat top leads. As a protection against moisture pieces of flat polyethylene taken from another piece of 300 -ohm Twin -Lead may be molded over the joint between conductors with the aid of an electric iron or soldering iron. Better bandwidth characteristics can be obtained with a folded dipole made of ribbon line if the two conductors of the ribbon line are shorted a distance of 0.82 (the velocity factor of ribbon line) of a free -space quarter wavelength from the center or feed point. This procedure is illustrated in figure 3A. An alternative arrangement for a Twin -Lead folded dipole is illustrated in figure 3B. This type of half -wave antenna system is convenient for use on the 3.5-Mc. band when the 116 to 132 foot distance required for a full half -wave is not quite available in a straight line, since the single -wire end pieces may be bent away or downward from the direction of the main section of the antenna. Figure 2K shows the basic type of 2 -wire doublet or folded dipole wherein the radiating section of the system is made up of standard antenna wire spaced by means of feeder spreaders. The feeder again is made of 300 ohm Twin -Lead since the feed -point impedance is approximately 300 ohms, the same as that of the Twin -Lead folded dipole. The folded -dipole type of antenna has the broadest response characteristic (greatest bandwidth) of any of the conventional halfwave antenna systems constructed of small wires or conductors. Hence such an antenna may be operated over the greatest frequency range without serious standing waves of any common half -wave antenna type. The increased bandwidth of the multi -wire doublet type of radiator, and the fact that the feed -point resistance is increased sever al www.americanradiohistory.com THE Antennas and Antenna Matching 424 Z e STUB -FED END -FED HERTZ EPP RADIO O A - 300 -600 LINE 11 END -FED TYPES r-O 0.95 A/2-1 --0.95 A/2--of- TUNED DOUBLET HALF-WAVE 300-600 O11M STUB- FED LINE 1 OPEN QUARTER-WAVE STUB- FED SHORTED 300-60011 LINE -0 .95 O A/2 r-- -+{ O T FOUR -WIRE 0-FEO 0.95 9/2 0.95 9/2 --{ 15011 TWINLEAD 0.193 OF FREE TWIN LEAD SPACE WAVELENGTH OR FED 4 0.77 OF LINE -FED 9/4 W---0.94 5/2{ 3000 IN CENTER O 2 -WIRE DOUBLET DOUBLET ANY LENGTH ANY LENGTH -0.95A/2-- FOR DELTA f~--0.95 A/2 DIMENSIONS SEE CHAP 19 DELTA MATCHED 300 OHM TWINLEAD 300 OHM TWINLEAD -+I ti 2 -OR FEEDER SPREADERS OR `FOLDED DIPOLE OLOEDDIPOIE - -0.95 5/2 6 TWINLEAD LOW SIDE OPENED n TWINLEAD ANY LENGTH 0.94 5/2 Y1/4 O TWINLEAD 300 60011 LINE 6000 LINE -41 600 OHM LINE ANY LENGTH r--095 A/2--.{ 14% OF 0 TOTAL LENGTH CO STANDARD -AA FED DOUBLET 750 Figure 2 ALTERNATIVE L N 14 WIRE TWINLEAD ANY LENGTH CENTER -FED TYPES, www.americanradiohistory.com METHODS OF FEEDING A HALF -WAVE DIPOLE Center -Fed Antennas TI ANT NUMBER OF HALF -WAVES FROM TRANSMITTER r ANY EVEN NUMBER OF QUARTER -WAVES f TV V I NIGH CAPACITANCE /1 T T LOW CAPACITANCE 423 retuning the feeders. The overall efficiency of the zepp antenna system is not quite as high for long feeder lengths as for some of the antenna systems which employ non -resonant transmission lines, but where space is limited and where operation on more than one band is desired, the zepp has some decided advantages. As the radiating portion of the zepp antenna system must always be some multiple of a half wave long, there is always high voltage present at the point where the live zepp feeder attaches to the end of the radiating portion of the antenna. Thus, this type of zepp antenna system is voltage led. I Stub -Fed Zepp- Figure THE END -FED HERTZ ANTENNA Showing the manner in which an end-fed Hertz antenna may be fed through a low -impedance line and low -pass filter by using a resonant tank circuit as at (A), or through the use of a reverse- connected pi network as at (B). Type Radiator 1 Some harmonic -attenuating provision (in addition to the usual low -pass TVI filter) must be included in the coupling system, as an end fed antenna itself offers no discrimination against harmonics, either odd or even. The end -fed Hertz antenna has rather high losses unless at least three -quarters of the radiator can be placed outside the operating room and in the clear. As there is r -f voltage at the point where the antenna enters the operating room, the insulation at that point should be several times as effective as the insulation commonly used with low- voltage feeder systems. This antenna can be operated on all of its higher harmonics with good efficiency, and can be operated at half frequency against ground as a quarter -wave Marconi. As the frequency of an antenna is raised slightly when it is bent anywhere except at a voltage or current loop, an end -fed Hertz antenna usually is a few per cent longer than a straight half-wave doublet for the same frequency, because, ordinarily, it is impractical to bring a wire in to the transmitter without making several bends. The zeppelin or zepp anterma system, illustrated in figure 2A is very convenient when it is desired to operate a single radiating wire on a number of harmonically reThe Zepp Antenna System lated frequencies. The zepp antenna system is easy to tune, and can be used on several bands by merely a non -resonant Figure 2C shows a modificaLion of the zepp -type antenna system to allow the use of transmission line between the radiating portion of the antenna and the transmitter. The zepp portion of the antenna is resonated as a quarter -wave stub and the non resonant feeders are connected to the stub at a point where standing waves on the feeder are minimized. The procedure for making these adjustments is described in detail in Section 22 -8 This type of antenna system is quite satisfactory when it is necessary physically to end feed the antenna, but where it is necessary also to use non -resonant feeder between the transmitter and the radiating system. 22 -2 Wave Center -Fed Half Horizontal Antennas The center feeding of a half -wave antenna system is usually to be desired over an end fed system since the center-fed system is inherently balanced to ground and is therefore less likely to be troubled by feeder radiation. A number of center -fed systems are illustrated in figure 2. The Tuned The current -fed do u b l e t with spaced feeders, sometimes called a center -fed zepp, is an inherently balanced system if the two legs of the radiator are electrically equal. This fact holds true regardless of the frequency, or of the harmonic, on which the system is operated. The system can successfully be operated over a wide range of frequencies if the system as a whole (both tuned feeders and the center -fed flat top) can be resonated to the operating frequency. It is usually possible to tune such an antenna system to resonance with the aid of a tapped coil and a tuning caDoublet pacitor that can optionally be placed either www.americanradiohistory.com CHAPTER TWENTY -TWO Antennas and Antenna Matching Antennas for the lower frequency portion of the h -f spectrum (perhaps from 1.8 to 7.0 Mc.), and temporary or limited use antennas for the upper portion of the h -f range, usually are of a relatively simple type in which directivity is not a prime consideration. Also, it often is desirable, in amateur work, that a single antenna system be capable of operation at least on the 3.5 -Mc. and 7.0 -Mc. range, and preferably on other frequency ranges. Consequently, the first portion of this chapter will be devoted to a discussion of such antenna systems. The latter portion of the chapter is devoted to the general problem of matching the antenna transmission line to antenna systems of the fixed type. Matching the antenna transmission line to the rotatable directive array is discussed in Chapter Twenty -five. 22 -1 End -Fed Half -Wave Horizontal Antennas Usually a high- frequency doublet is mounted as high and as much in the clear as possible, for obvious reasons. However, it is sometimes justifiable to bring part of the radiating system directly to the transmitter, feeding the antenna without benefit of a transmission line. This is permissible when (1) there is insufficient room to erect a 75- or 80 -meter horizontal dipole and feed line, (2) when a long wire is also to be operated on one of the higher frequency bands on a harmonic. In either case, it is usually possible to get the main portion of the antenna in the clear because of its length. This means that the power lost by bringing the antenna directly to the transmitter is relatively small. Even so, it is not best practice to bring the high -voltage end of an antenna into the operating room because of the increased difficulty in eliminating BC! and TVI. For this reason one should dispense with a feed line in conjunction with a Hertz antenna only as a last resort. The end -fed antenna has no form of transmission line to couple it to the transmitter, but brings the radiating portion of the antenna right down to the transmitter, where some form of coupling system is used to transfer energy to the anEnd -Fed The half -wave horizontal dipole is the most common and the most practical antenna for the 3.5 -Mc. and 7 -Mc. amateur bands. The form of the dipole, and the manner in which it is fed are capable of a large number of variations. Figure 2 shows a number of practicable forms of the simple dipole antenna along with methods of feed. Antennas tenna. Figure 1 shows two common methods of feeding the Fuchs antenna or end -fed Hertz. 422 www.americanradiohistory.com HANDBOOK Tuned amplitude, in turn, depends upon the mismatch at the line termination. A line of no. 12 wire, spaced 6 inches with good ceramic or plastic spreaders, has a surge impedance of approximately 600 ohms, and makes an excellent tuned feeder for feeding anything between 60 and 6000 ohms (at frequencies below 30 Mc.). If used to feed a load of higher or lower impedance than this, the standing waves become great enough in amplitude that some loss will occur unless the feeder is kept short. At frequencies above 30 Mc., the spacing becomes an appreciable fraction of a wavelength, and radiation from the line no longer is negligible. Hence, coaxial line or close- spaced parallel wire line is recommended for v -h -f work. If a transmission line is not perfectly matched, it should be made resonant, even though the amplitude of the standing waves (voltage variation) is not particularly great. This prevents reactance from being coupled into the final amplifier. A feed system having moderate standing waves may be made to present a non reactive load to the amplifier either by tuning or by pruning the feeders to approximate resonance. Usually it is preferable with tuned feeders to have a current loop (voltage minimum) at the transmitter end of the line. This means that when voltage- feeding an antenna, the tuned feeders should be made an odd number of quarter wavelengths long, and when current -feeding an antenna, the feeders should be made an even number of quarter wavelengths long. Actually, the feeders are made about 10 per cent of a quarter wave longer than the calculated value (the value given in the tables) when they are to be series tuned to resonance by means of a capacitor, instead of being trimmed and pruned to resonance. When tuned feeders are used to feed an antenna on more than one band, it is necessary to compromise and make provision for both series and parallel tuning, inasmuch as it is impossible to cut a feeder to a length that will be optimum for several bands. If a voltage loop appears at the transmitter end of the line on certain bands, parallel tuning of the feeders will be required in order to get a transfer of energy. It is impossible to transfer energy by inductive coupling unless current is flow- ing. This is effected at a voltage loop by the Lines 421 presence of the resonant tank circuit formed by parallel tuning of the antenna' coil. 21 -12 Line Discontinuities In the previous discussion we have assumed transmission line which was uniform throughout its length. In actual practice, this is usually not the case. Whenever there is any sudden change in the characteristic impedance of the line, partial reflection will occur at the point of discontinuity. Some of the energy will be transmitted and some reflected, which is essentially the same as having some of the energy absorbed and some reflected in so far as the effect upon the line from the generator to that point is concerned. The discontinuity can by ascribed a reflection coefficient just as in the case of an unmatched load. In a simple case, such as a finite length of uniform line having a characteristic impedance of 500 ohms feeding into an infinite length of uniform line having a characteristic impedance of 100 ohms, the behavior is easily predicted. The infinite 100 ohm lin& will have no standing waves and will accept the same power from the 500 ohm line as would a 100 ohm resistor, and the rest of the energy will be reflected at the discontinuity to produce standing waves from there back to the generator. However, in the case of a complex discontinuity placed at an odd distance down a line terminated in a complex impedance, the picture becomes complicated, especially when the discontinuity is neither sudden nor gradual, but intermediate between the two. This is the usual case with amateur lines that must be erected around buildings and trees. In any case, when a discontinuity exists somewhere on a line and is not a smooth, gradual change embracing several wavelengths, it is not possible to avoid standing waves throughout the entire length of the line. If the discontinuity is sharp enough and is great enough to be significant, standing waves must exist on one side of the discontinuity, and may exist on both sides in many cases. a www.americanradiohistory.com 420 line fed by a transmitter. It is the reflection from the antenna end which starts waves moving back toward the transmitter end. When waves moving in both directions along a conductor meet, standing waves are set up. well- constructed open wire line has acceptably Parallel -Wire Lines low losses when its length is less than about two wavelengths even when the voltage standing -wave ratio is as high as 10 to 1. A transmission line constructed of ribbon or tubular line, however, should have the standing -wave ratio kept down to not more than about 3 to 1 both to reduce power loss and because the energy dissipation on the line will be localized, causing overheating of the line at the points of maximum current. Because moderate standing waves can be tolerated on open -wire lines without much loss, a standing -wave ratio of 2/1 or 3/1 is considered acceptable with this type of line, even when used in an untuned system. Strictly speaking, a line is untuned, or non -resonant, only when it is perfectly flat, with a standing wave ratio of 1 (no standing waves). However, some mismatch can be tolerated with open-wire untuned lines, so long as the reactance is not objectionable, or is eliminated by cutting the line to approximately resonant length. Semi -Resonant 21 -11 THE Radiation, Propagation and Lines t Zo 1;) 1.0 o A Tuned or Resonant Lines If a transmission line is terminated in its characteristic surge impedance, there will be no reflection at the end of the line, and the current and voltage distribution will be uniform along the line. If the end of the line is either open- circuited or short -circuited, the reflection at the end of the line will be 100 per cent, and standing waves of very great amplitude will appear on the line. There will still be practically no radiation from the line if it is closely spaced, but voltage nodes will be found every half wavelength, the voltage loops corresponding to current nodes (figure 23). If the line is terminated in some value of resistance other than the characteristic surge impedance, there will be some reflection, the amount being determined by the amount of mismatch. With reflection, there will be standing waves (excursions of current and voltage) along the line, though not to the same extent as with an open- circuited or short- circuited line. The current and voltage loops will occur at the same points along the line as with the open or short- circuited line, and as the terminating impedance is made to approach the characteristic impedance of the line, the cur- RADIO swR t.o ZL. SWR = 1.5 Zo ZL. +.s on 0.e1 Zo 1.s o SWR 3.0 ZL 3.0 OR 0.31 ZO SWRo ZLooRo Figure 23 STANDING WAVES ON A TRANSMISSION LINE As shown at (A), the voltage and current are constant on a transmission line which is terminated in its characteristic impedance, assuming that losses are small enough so that they may be neglected. (B) shows the variation in current or in voltage on a line terminated in a load with a reflection coefficient of 0.2 so that a standing wave ratio of 1.5 to I is set up. At (C) the reflection coefficient has been increased to 0.5, with the formation of a 3 to 1 standing -wove ratio on the line. At (D) the line has been terminated in a load which has a reflection coefficient of I.0 (short, open circuit, or a pure reactance) so that all the energy is reflected with the formation of an infinite standing wave ratio. rent and voltage along the line will become more uniform. The foregoing assumes, of course, a purely resistive (non -reactive) load. If the load is reactive, standing waves also will be formed. But with a reactive load the nodes will occur at different locations from the node locations encountered with wrong- value resistive termination. A well built 500- to 600 -ohm transmission line may be used as a resonant feeder for lengths up to several hundred feet with very low loss, so long as the amplitude of the standing waves (ratio of maximum to minimum voltage along the line) is not too great. The www.americanradiohistory.com THE RADIO Transmission ribbon and tubular configuration, with characteristic impedance values from 75 to 300 ohms. Receiving types, and transmitting types for power levels up to one kilowatt in the h -f range, are listed with their pertinent characteristics, in the table of figure 21. 419 Lines 204 Zo7SSLOGp 170 COAXIAL OR CONCENTRIC LINE us Coaxial Line Several types of coaxial cable have come into wide use for feeding power to an antenna system. A cross sectional view of a coaxial cable (sometimes called concentric cable or line) is shown in figure 22. As in the parallel -wire line, the power lost in a properly terminated coaxial line is the sum of the effective resistance losses along the length of the cable and the dielectric losses between the two conductors. Of the two losses, the effective resistance loss is the greater; since it is largely due to the skin effect, the line loss (all other conditions the same) will increase directly as the square root of the frequency. Figure 22 shows that, instead of having two conductors running side by side, one of the conductors is placed inside of the other. Since the outside conductor completely shields the inner one, no radiation takes place. The conductors may both be tubes, one within the other; the line may consist of a solid wire within a tube, or it may consist of a stranded or solid inner conductor with the outer conductor made up of one or two wraps of copper shielding braid. In the type of cable most popular for military and non -commercial use the inner conductor consists of a heavy stranded wire, the outer conductor consists of a braid of copper wire, and the inner conductor is supported within the outer by means of a semi -solid dielectric of exceedingly low loss characteristics called polyethylene. The Army -Navy designation on one size of this cable suitable for power levels up to one kilowatt at frequencies as high as 30 Mc. is AN /RG -8 /U. The outside diameter of this type of cable is approximately one -half inch. The characteristic impedance of this cable type is 52 ohms, but other similar types of greater and smaller power-handling capacity are available in impedances of 52, 75, and 95 ohms. When using solid dielectric coaxial cable it is necessary that precautions be taken to insure that moisture cannot enter the line. If the better grade of connectors manufactured for the line are employed as terminations, this condition is automatically satisfied. If connectors are not used, it is necessary that some type of moisture-proof sealing compound be applied to the end of the cable where it will be exposed to the weather. Nearby metallic objects cause no loss, and coaxial cable may be run up air ducts or ele- loo Di. INSIDE 70 DIAMETER OF OUTER CONDUCTOR s2 D= OUTSIDE DIAMETER OF INNER CONDUCTOR 30 o 2.81 3.21 5 7 to iS 30 RATIO OF DIAMETERS Figure 22 CHARACTERISTIC IMPEDANCE OF AIR FILLED COAXIAL LINES If the filling of the line is o dielectric material other than air, the characteristic impedance of the line will be reduced by a factor proportional to the square -root of the dielectric constant of the material used as a dielectric within the line. vator shafts, inside walls, or through metal conduit. Insulation troubles can be forgotten. The coaxial cable may be buried in the ground or suspended above ground. Standing Waves Standing waves on a transmission line always are the result of the reflection of energy. The only significant reflection which takes place in a normal installation is that at the load end of the line. But reflection can take place from discontinuities in the line, such as caused by insulators, bends, or metallic objects adjacent to an unshielded line. When a uniform transmission line is terminated in an impedance equal to its surge impedance, reflection of energy does not occur, and no standing waves are present. When the load termination is exactly the same as the line impedance, it simply means that the load takes energy from the line just as fast as the line delivers it, no slower and no faster. Thus, for proper operation of an untuned line (with standing waves eliminated), some form of impedance- matching arrangement must be used between the transmission line and the antenna, so that the radiation resistance of the antenna is reflected back into the line as a nonreactive impedance equal to the line impedance. The termination at the antenna end is the only critical characteristic about the untuned www.americanradiohistory.com 418 RADIO THE Radiation, Propagation and Lines CHARACTERISTICS OF COMMON TRANSMISSION LINES ATTENUATION db/ VELOCITYUUFD 00 FEET vswR =1.0 1- 30 Mc 100 MC 300 MC OPEN WIRE LINE, COPPER. N' 12 RIBBON LINE, REC.TYPE, 300 OHMS. (7/2e 0.3 0.6 0.86 2.2 5.3 O.q8 -099 0.62 ., 1-271) FT Ny 6 RIBBON LINE, TRANS. TYPE. 300 OHMS. - - - - - - - - TUBULAR "TWIN -LEADTRANS. TYPE, 7/160.D. 0.65 (AMPHENOL 14 -076) 2 3 5.4 RIBBON LIKE, RECEIVE TYPE, ISO OHMS. 2 7 6 1 1 O REMARKS PER CONDUCTORS) TUBULAR "TWIN-LEAD" REC TYPE. 300 OHMS, S /16.0.0., (AMPHENOL TYPE 0.15 ACTOR V BASED UPON 4" SPACING BELOW 50 MC ; 2- SPACING ABOVE 50 MC. RADIATION LOSSES INCLUDED. CLEAN, LOW LOSS CERAMIC INSULATION ASSUMED RADIATION HIGH ABOVE 150 MC FOR CLEAN. DRY LINE. wET WEATHER PERFORMANCE RATHER POOR BEST LINE IS SLIGHTLY CONVEX. AVOID LINE THAT HAS CONCAVE DIELECTRIC SUITABLE FOR LOW POWER TRANSMITTING APPLICATIONS. LOSSES INCREASE AS LINE WEATHERS. HANDLES 400 WATTS AT 30 MC. IF VSWR IS LOW. CHARACTERISTICS SIMILAR TO RECEIVING TYPE RIBBON LINE EXCEPT FOR MUCH BETTER wET WEATHER PERFORMANCE. CHARACTERISTICS VARY SOMEWHAT WITH MANUFACTURER. BUT APPROXIMATE THOSE OF RECEIVING TYPE RIBBON EXCEPT FOR GREATER POWER HANDLING CAPABILITY AND SLIGHTLY BETTER WET WEATHER PERFORMANCE. 0.79 8.1 FOR USE WHERE RECEIVING TYPE TUBULAR -TWIN -LEAD DOES NOT HAVE SUMCIENT POWER HANDLING CAPABILITY. WILL HANDLE / KW AT 30 MC. F VSWR IS LOW. 0 77 V' 10 USEFUL FOR QUARTER WAVE MATCHING SECTIONS. I AS A NO LONGER WIDELY USED LINE. USEFUL MAINLY IN THE H -F RANGE BECAUSE OF EXCESSIVE LOSSES AT V -H -F AND U-H-F. LESS AFFECTED BY WEATHER THAN 300 OHM_RIBBON. VERY SATISFACTORY FOR TRANSMITTING APPLICATIONS BELOW 30 MC. AT KW. NOT SIGNIFICANTLY AFFECTED BY WET WEATHER. POWERS UP TO RIBBON LINE, RECEIVE. TYPE, 75 OHMS. 2 5 O 11.0 o.BB' 19 V' RIBBON LINE, TRANS. TYPE, 75 OHMS. 1.5 3.9. 6.0 0.71Y f RG-6/U COAX (52 OHMS) 1.0 2.1 4.2 0.88 29.5 WILL HANDLE 2 KW AT 4O MC. IF VSWR RG-11 /U COAX (75 OHMS) 0.94 I 9 3.6 0.88 20.5 WILL HANDLE 1. RG -17 /U COAX (520HMS) 0.38 0.85 1.8 0.66 29.5 WILL HANDLE 7 1.95 4.1 8.0 0.66 28.5 WILL HANDLE 430 WATTS AT 30 MC. IF VSWR IS LOW. 0.2000. RG -58/U COAX (53 OHMS) O 6`t 1 KW AT IS LOW. 30 MC. IF VSWR 0.. IS LOW. 8 KW. AT 30 MC. IF VSWR IS LOW. 0 O.D. 7/21 CONDUCTOR. 4 "0.0. 7/28 CONDUCTOR. 087" OD. 0.19" 0.24" O.D. DIA. CONDUCTOR N 20 CONDUCTOR. N 22 CONDUCTOR. RG-S9 /U COAX (73 OHMS) 1.9 3.8 7.0 0.66 21 WILL HANDLE 680 WATTS AT 30 MC. IF VSWR (720HMS) 2.0 4.0 7.0 0.66 22 COMMERCIAL VERSION OF RG-59/U FOR LESS EXACTING APPLICATIONS. EXPENSIVE. FOR SHIELDED, BALANCED -TO- GROUND APPLICATIONS. VERY LOW NOISE PICK UP. 0.4" 0.D. TV -59 COAX RG -22/U SHIELDED PAIR (95 OHMS) K -I11 SHIELDED PAIR (300 OHMS) 1.7 3.0 5.5 0.66 18 2.0 3.5 6.1 - 4 0 APPROXIMATE. EXACT FIGURE VARIES SLIGHTLY WITH MANUFACTURER 2S = 276 1og10d Where: S is the exact distance between wire centers in some convenient unit of measurement, and d is the diameter of the wire measured in the same units as the wire spacing, S. 2S Since LESS DESIGNED FOR TV LEAD -IN IN NOISY LOCATIONS. LOSSES HIGHER THAN REGULAR 300 OHM RIBBON, BUT DO NOT INCREASE AS MUCH FROM WEATHERING FIGURE Z. IS LOW. 21 Surge impedance values of less than 200 ohms are seldom used in the open -type two wire line, and, even at this rather high value of Z. the wire spacing S is uncomfortably close, being only 2.7 times the wire diameter d. Figure 20 gives in graphical form the surge impedance of practicable two -wire lines. The chart is self-explanatory, and is sufficiently accurate for practical purposes. Instead of using spacer insulators placed periodically along the transmission line it is possible to mold the line conductors into a ribbon or tube of flexible low -loss dielectric material. Such line, with polyethylene dielectric, is used in enormous quantities as the lead -in transmission line for FM and TV receivers. The line is available from several manufacturers in the Ribbon and expresses a ratio only, the units d of measurement may be centimeters, millimeters, or inches. This makes no difference in the answer, so long as the substituted values for S and d are in the same units. The equation is accurate so long as the wire spacing is relatively large as compared to the wire diameter. Tubular Transmission Line www.americanradiohistory.com HANDBOOK Transmission ever, mechanical or electrical considerations often make one type of transmission line better adapted for use to feed a particular type of antenna than any other type. Transmission lines for carrying r -f energy are of two general types: non -resonant and resonant. A non -resonant transmission line is one on which a successful effort has been made to eliminate reflections from the termination (the antenna in the transmitting case and the receiver for a receiving antenna) and hence one on which standing waves do not exist or are relatively small in magnitude. A resonant line, on the other hand, is a transmission line on which standing waves of appreciable magnitude do appear, either through inability to match the characteristic impedance of the line to the termination or through intentional design. The principal types of transmission line in use or available at this time include the open wire line (two -wire and four -wire types), two wire solid -dielectric line ( "Twin- Lead" and similar ribbon or tubular types), two -wire polyethylene- filled shielded line, coaxial line of the solid -dielectric, beaded, stub-supported, or pressurized type, rectangular and cylindrical wave guide, and the single -wire feeder operated against ground. The significant characteristics of the more popular types of transmission line available at this time are given in the chart of figure 21. 21 -10 Non -Resonant Transmission Lines A non -resonant or untuned transmission line is a line with negligible standing waves. Hence, a non -resonant line is a line carrying r -f power only in one direction -from the source of energy to the load. Physically, the line itself should be identical throughout its length. There will be a smooth distribution of voltage and current throughout its length, both tapering off very slightly towards the load end of the line as a result of line losses. The attenuation (loss) in certain types of untuned lines can be kept very low for line lengths up to several thousand feet. In other types, particularly where the dielectric is not air (such as in the twisted pair line), the losses may become excessive at the higher frequencies, unless the line is relatively short. Transmission -Line All transmission lines have distributed inductance, capacitance and resist ance. Neglecting the resistance, as it is of minor importance in short lines, it is found Impedance Lines 417 1111111111ESSMINI%s's_í.s %%M III / IIM/11111Ï11 ILW /Oií%E'/_Er1111111111 .-- iiilMEM ;mt=1 s . io is 3 s INCHES. CENTER TO CENTER 111111111 + w w u Figure 20 CHARACTERISTIC IMPEDANCE OF TYPICAL TWO -WIRE OPEN LINES that the inductance and capacitance per unit length determine the characteristic or surge impedance of the line. Thus, the surge impedance depends upon the nature and spacing of the conductors, and the dielectric separating them. Speaking in electrical terms, the characteristic impedance of a transmission line is simply the ratio of the voltage across the line to the current which is flowing, the same as is the case with a simple resistor: Z. = E /1. Also, in a substantially loss -less line (one whose attenuation per wavelength is small) the energy stored in the line will be equally divided between the capacitive field and the inductive field which serve to propagate the energy along the line. Hence the characteristic impedance of a line maybe expressed as: Z. Two -Wire Open Line = N/ L/C. two -wire transmission system is easy to construct. Its surge impedance can be calculated quite easily, and when properly adjusted and balanced to ground, with a conductor spacing which is negligible in terms of the wavelength of the signal carried, undesirable feeder radiation is minimized; the current flow in the adjacent wires is in opposite directions, and the magnetic fields of the two wires are in opposition to each other. When a two -wire line is terminated with the equivalent of a pure resistance equal to the characteristic impedance of the line, the line becomes a non resonant line. Expressed in physical terms, the characteristic impedance of a two -wire open line is A equal to: www.americanradiohistory.com 416 THE Radiation, Propagation and Lines dent, particularly a "flutter fade" and a characteristic "hollow" or echo effect. Deviations from a great circle path are especially noticeable in the case of great circle paths which cross or pass near the auroral zones, because in such cases there often is complete or nearly complete absorption of the direct sky wave, leaving off -path scattered reflections the only mechanism of propagation. Under such conditions the predominant wave will appear to arrive from a direction closer to the equator, and the signal will be noticeably if not considerably weaker than a direct sky wave which is received under favorable conditions. Irregular reflection of radio waves from "scattering patches" is divided into two categories: "short scatter" and "long scatter ". Short scatter is the scattering that occurs when a radio wave first reaches the scattering patches or media. Ordinarily it is of no particular benefit, as in most cases it only serves to fill in the inner portion of the skip zone with a weak, distorted signal. Long scatter occurs when a wave has been refracted from the F2 layer and strikes scattering patches or media on the way down. When the skip distance exceeds several hundred miles, long scatter is primarily responsible for reception within the skip zone, particularly the outer portion of the skip zone. Distortion is much less severe than in the case of short scatter, and while the signal is likewise weak, i t sometimes can be utilized for satisfactory communication. During a severe ionosphere disturbance in the north auroral zone, it sometimes is possible to maintain communication between the Eastern United States and Northern Europe by the following mechanism: That portion of the energy which is radiated in the direction of the great circle path is completely absorbed upon reaching the auroral zone. However, the portion of the wave leaving the United States in a southeasterly direction is refracted downward from the F2 layer and encounters scattering patches or media on its downward trip at a distance of approximately 2000 miles from the transmitter. There it is reflected by "long scatter" in all directions, this scattering region acting like an isotropic radiator fed with a very small fraction of the original transmitter power. The great circle path from this southerly point to northern Europe does not encounter unfavorable ionosphere conditions, and the wave is propagated the rest of the trip as though it had been radiated from the scattering region. Another type of scatter is produced when a sky wave strikes certain areas of the earth. Upon striking a comparatively smooth surface such as the sea, there is little scattering, the wave being shot up again by what could be RADIO considered specular or mirror reflection. But upon striking a mountain range, for instance, the reradiation or reflected energy is scattered, some of it being directed back towards the transmitter, thus providing another mechanism for producing a signal within the skip zone. strikes the earth's atmosphere, a cylindrical region of free electrons is formed at approximately the height of the E layer. This slender ionized column is quite long, and when first formed is sufficiently dense to reflect radio waves back to earth most readily, including v -h -f waves which are not ordinarily returned by the F= layer. The effect of a single meteor, of normal size, shows up as a sudden "burst" of signal of short duration at points not ordinarily reached by the transmitter. After a period of from 10 to 40 seconds, recombination and diffusion have progressed to the point where the effect of a single fairly large meteor is not perceptible. However, there are many small meteors impinging upon earth's atmosphere every minute, and the aggregate effect of their transient ionized trails, including the small amount of residual ionization that exists for several minutes after the original flash but is too weak and dispersed to prolong a "burst ", is believed to contribute to the existence of the "nighttime E" layer, and perhaps also to sporadic E patches. While there are many of these very small meteors striking the earth's atmosphere every minute, meteors of normal size (sufficiently large to produce individual "bursts ") do not strike nearly so frequently except during some of the comparatively rare meteor "showers ". Metéors and When a meteor "Bursts" During one of these displays a "quivering" ionized layer is produced which is intense enough to return signals in the lower v -h -f range with good strength, but with a type of "flutter" distortion which is characteristic of this type of propagation. 21 -9 Transmission Lines For many reasons it is desirable to place an antenna or radiating system as high and in the clear as is physically possible, utilizing some form of nonradiating transmission line to carry energy with as little loss as possible from the transmitter to the radiating antenna, and conversely from the antenna to the re- ceiver. There are many different types of transmission lines and, generally speaking, practically any type of transmission line or feeder system may be used with any type of antenna. How- HANDBOOK 11 -Year Sunspot Cycle 415 225 200 i' 175 150 é 125 50 '5 ó; C ; I , 2 ,-` , . 3 25 YEAR o 48 ttt ri 100 75 , 50 52 54 58 Figure 56 80 62 64 86 18 THE YEARLY TREND OF THE SUNSPOT CYCLE. RADIO CONDITIONS IN GENERAL WILL DETERIORATE DURING 19601965 AS THE CYCLE DECLINES. zon, the farther away will the wave return to earth, and the greater the skip distance. The wave can be reflected back up into the ionosphere by the earth, and then be reflected back down again, causing a second skip distance area. The drawing of figure 19 shows the multiple reflections possible. When the receiver receives signals which have traveled over more than one path between transmitter and receiver, the signal impulses will not all arrive at the same instant, as they do not all travel the same distance. When two or more signals arrive in the same phase at the receiving antenna, the resulting signal in the receiver will be quite strong. On the other hand, if the signals arrive 180° out of phase, so they tend to cancel each other, the received signal will drop -perhaps to zero if perfect cancellation occurs. This explains why high -frequency signals are subject to fading. Fading can be greatly reduced on the high frequencies by using a transmitting antenna with sharp vertical directivity, thus cutting down the number of possible paths of signal arrival. A receiving antenna with similar characteristics (sharp vertical directivity) will further reduce fading. It is desirable, when using antennas with sharp vertical directivity, to use the lowest vertical angle consistent with good signal strength for the frequency used. Scattered Reflections Scattered reflections are random, diffused, substantially isotropic reflections which are partly re- TRANSMITTER Figure 19 IONOSPHERE -REFLECTION WAVE PATHS Showing typical ionosphere- reflection wave paths during daylight hours when ionization density is such that frequencies as high as 28 Mc. will he returned to earth. The distance between ground -wave range and that range where the ionosphere -reflected wove of a specific frequency first will be returned to earth is called the skip distance. sponsible for reception within the skip zone, and for reception of signals from directions off the great circle path. In a heavy fog or mist, it is difficult to see the road at night because of the bright glare caused by scattered reflection of the headlight beam by the minute droplets. In fact, the road directly to the side of the car will be weakly illuminated under these conditions, whereas it would riot on a clear night (assuming flat, open country). This is a good example of propagation of waves by scattered reflections into a zone which otherwise would not be illuminated. Scattering occurs in the ionosphere at all times, because of irregularities in the medium (which result in "patches" corresponding to the water droplets) and because of random phase radiation due to the collision or recombination of free electrons. However, the nature of the scattering varies widely with time, in a random fashion. Scattering is particularly prevalent in the f: region, but scattered reflections may occur at any height, even well out beyond the virtual height of the /-'2 layer. There is no "critical frequency" or "lowest perforating frequency" involved in the scattering mechanism, though the intensity of the scattered reflections due to typical scattering in the F. region of the ionosphere decreases with frequency. CChen the received signal is due primarily to scattered reflections, as is the case in the skip zone or where the great circle path does not provide a direct sky wave (due to low critical or perforation frequency, or to an ionosphere storm) very bad distortion will be evi- www.americanradiohistory.com 10 34 32 30 WINTER 2e SUNSPOT MAXIMUM 26 l' 1 24 22 ï 20 U 16 w 14 a 12 ù Z rc a T Radiation, Propagation and Lines 414 -.1,, III SUMMER SUNSPOTMINIMUM - 4 2 0 2 6 e 10 12 14 16 16 ii' 20 22 24 LOCAL TIME Figure 17 TYPICAL CURVES SHOWING CHANGE M.U.F. IN AT MINIMUM MAXIMUM AND POINTS IN SUNSPOT CYCLE The m.u.f. often drops to frequencies below the early morning hours. The high m.u.f. in the middle of the day is brought about by reflection from the F2 layer. M.u.f. data is published periodically in the magazines devoted to amateur work, and the m.u.f. can be calculated with the aid of Basic Radio Propagation Predictions, CRPL -D, published monthly by the Government Printing Office, Washington, D.C. 10 Mc. in The optimum working frequency for any particular Frequency direction and distance is usually about 15 per cent less than the m.u.f. for contact with that particular location. The absorption by the ionosphere becomes greater and greater as the operating frequency is progressively lowered below the m.u.f. It is this condition which causes signals to increase tremendously in strength on the 14 -Mc. and 28 -Mc. bands just before the signals drop completely out. At the time when the signals are greatest in amplitude the operating frequency is equal to the m.u.f. Then as the signals drop out the m.u.f. has become lower than the operating frequency. Absorption and Optimum Working The shortest distance from a transmitting location at which signals reflected from the ionosphere can be returned to the earth is called the skip distance. As was mentioned above under Critical Frequency there is no skip distance for a frequency below the critical frequency of the Skip Distance E R A D I O most highly ionized layer of the ionosphere at the time of transmission. However, the skip distance is always present on the 14 -Mc. band and is almost always present on the 3.5 -Mc. and 7 -Mc. bands at night. The actual measure of the skip distance is the distance between the point where the ground wave falls to zero and the point where the sky wave begins to return to earth. This distance may vary from 40 to 50 miles on the 3.5-Mc. band to thousands of miles on the 28-Mc. band. Occasional patches of extremely high ionization density appear at intervals throughout the year at a height approximately equal to that of the F layer. These patches, called the sporadic-F. layer may be very small or may be up to several hundred miles in extent. The critical frequency of the sporadic-F layer may be greater than twice that of the normal ionosphere layers which exist at the The Sporadic -E Loyer e e H same time. It is this sporadic -E condition which provides "short- skip" contacts from 400 to perhaps 1200 miles on the 28 -Mc. band in the evening. It is also the sporadic-E condition which provides the more common type of "band opening" experienced on the 50 -Mc. band when very loud signals are received from stations from 400 to 1200 miles distant. Cycles in The ionization density of the ionosphere is deterIonosphere Activity mined by the amount of radiation (probably ultra violet) which is being received from the sun. Consequently, ionosphere activity is a function of the amount of radiation of the proper character being emitted by the sun and is also a function of the relative aspect of the regions in the vicinity of the location under discussion to the sun. There are four main cycles in ionosphere activity. These cycles are: the daily cycle which is brought about by the rotation of the earth, the 27 -day cycle which is caused by the rotation of the sun, the seasonal cycle which is caused by the movement of the earth in its orbit, and the 11-year cycle which is a cycle in sunspot activity. The effects of these cycles are superimposed insofar as ionosphere activity is concerned. Also, the cycles are subject to short term variations as a result of magnetic storms and similar terrestrial disturbances. The most recent minimum of the 11 -year sunspot cycle occurred during the winter of 19541955, and we are currently moving along the slope of a new cycle, the maximum of which occurred during 1958. The current cycle is pictured in figure 18. Fading The lower the angle of radiation of the wave, with respect to the hori- www.americanradiohistory.com HANDBOOK Ionospheric Propagation 200 virtual height of approximately 175 miles at night, and in the daytime it splits up into two layers, the upper one being called the F, layer and the lower being called the F, layer. The height of the F2 layer during daylight hours is normally about 250 miles on the average and the F, layer often has a height of as low as 140 miles. It is the F2 layer which supports all nighttime dx communication and nearly all daytime dx propagation. a F2 130 Ft too MID DAY E D F2 Below the F2 layer is another layer, called the E layer, which is of importance in daytime communication over moderate distances in the frequency range between 3 and 8 Mc. This layer has an almost constant height at about 70 miles. Since the re- combination time of the ions at this height is rather short, the E layer disappears almost completely a short time after local sunset. The MIDNIGHT too 50 IONIZATION DENSITY -a Figure 16 IONIZATION DENSITY IN THE IONOSPHERE Showing typical ionization density of the ionosphere in mid -summer. Note that the Ft and D layers disappear at night, and that the density of the E layer falls to such a low value that it is ineffective. which the sky wave can undergo depends upits frequency, and the amount of ionization in the ionosphere, which is in turn dependent upon radiation from the sun. The sun increases the density of the ionosphere layers (figure 16) and lowers their effective height. For this reason, the ionosphere acts very differently at different times of day, and at different times of the year. The higher the frequency of a radio wave, the farther it penetrates the ionosphere, and the less it tends to be bent back toward the earth. The lower the frequency, the more easily the waves are bent, and the less they penetrate the ionosphere. 160 -meter and 80-meter signals will usually be bent back to earth even when sent straight up, and may be considered as being reflected rather than refracted. As the frequency is raised beyond about 5,000 kc. (dependent upon the critical frequency of the ionosphere at the moment), it is found that waves transmitted at angles higher than a certain critical angle never return to earth. Thus, on the higher frequencies, it is necessary to confine radiation to low angles, since the high angle waves simply penetrate the ionosphere and are lost. on The F2 Layer 413 The higher of the two major reflection regions of the ionosphere is called the F, layer. This layer has E Layer Below the E layer at a height of about 35 miles is an absorbing layer, called the D layer, which exists in the middle of the day in the summertime. The layer also exists during midday in the winter time during periods of high solar activity, but the layer disappears completely at night. It is this layer which causes high absorption of signals in the medium and high- frequency range during the middle of the day. The D Layer Critical Frequency critical frequency of ionospheric layer is the highest frequency which will be reflected when the wave strikes the layer at vertical incidence. The critical frequency of the most highly ionized layer of the ionosphere may be as low as 2 Mc. at night and as high as 12 to 13 Mc. in the middle of the day. The critical frequency is directly of interest in that a skip distance zone will exist on all frequencies greater than the highest critical frequency at that time. The critical frequency is a measure of the density of ionization of the reflecting layers. The higher the critical frequency the greater the density of ionization. The an Maximum Usable The maximum usable /requency or m.u. /. is of great importance in long- distance communication since this frequency is the highest that can be used for communication between any two specified areas. The m.u.f. is the highest frequency at which a wave projected into space in a certain direction will be returned to earth in a specified region by ionospheric reflection. The m.u.f. is highest at noon or in the early afternoon and is highest in periods of greatest sunspot activity, often going to frequencies higher than 50 Mc. Frequency (figure 17). www.americanradiohistory.com THE Radiation, Propagation and Lines 412 T __ INVERSION DUCT T INVERSION AND DUCT REFRACTIVE INDEX Figure 15 ILLUSTRATING DUCT TYPES Showing two types of variation in refractive index with height which will give rise to the formation of a duct. An elevated duct is shown at (A), and o ground -based duct is shown at (B). Such ducts can propagate ground-wave signals far beyond their normal range. rise to the formation of a duct which can propagate waves with very little attenuation over great distances in a manner similar to the propagation of waves through a wave guide. Guided propagation through a duct in the atmosphere can give quite remarkable transmission conditions (figure 15). However, such ducts usually are formed only on an over water path. The depth of the duct over the water 's surface may be only 20 to 50 feet, or it may be 1000 feet deep or more. Ducts exhibit a low -frequency cutoff characteristic similar to a wave guide. The cutoff frequency is determined by depth of the duct and by the strength of the discontinuity in refractive index at the upper surface of the duct. The low est'frequency that can be propagated by such a duct seldom goes below 50 Mc., and usually will be greater than 100 Mc. even along the may give Pacific Coast. virtue of Stratospheric Communication Reflection stratospheric reflection can be by brought about during magnetic storms, aurora borealis displays, and during meteor showers. Dx communication during extensive meteor showers is characterized by frequent bursts of great signal strength followed by a rapid decline in strength of the received signal. The motion of the meteor forms an ionized trail of considerable extent which can bring about effective reflection of signals. However, the ionized region persists only for a matter of seconds so that a shower of meteors is necessary before communication becomes possible. The type of communication which is possible during visible displays of the aurora borealis RADIO and during magnetic storms has been called aurora-type dx. These conditions reach a maximum somewhat after the sunspot cycle peak, possibly because the spots on the sun are nearer to its equator (and more directly in line with the earth) in the latter part of the cycle. Ionospheric storms generally accompany magnetic storms. The normal layers of the ionosphere may be churned or broken up, making radio transmission over long distances difficult or impossible on high frequencies. Unusual conditions in the ionosphere sometimes modulate v -h-f waves so that a definite tone or noise modulation is noticed even on transmitters located only a few miles away. A pecularity of this type of auroral propagation of v -h -f signals in the northern hemisphere is that directional antennas usually must be pointed in a northerly direction for best results for transmission or reception, regardless of the direction of the other station being contacted. Distances out to 700 or 800 miles have been covered during magnetic storms, using 30 and 50 Mc. transmitters, with little evidence of any silent zone between the stations communicating with each other. Generally, voice -modulated transmissions are difficult or impossible due to the tone or noise modulation on the signal. Most of the communication of this type has taken place by c.w. or by tone modulated waves with a keyed carrier. 21-8 Ionospheric Propagation Propagation of radio waves for communication on frequencies between perhaps 3 and 30 Mc. is normally carried out by virtue of ionospheric reflection or refraction. Under conditions of abnormally high ionization in the ionosphere, communication has been known to have taken place by ionospheric reflection on frequencies higher than 50 Mc. The ionosphere consists of layers of ionized gas located above the stratosphere, and extending up to possibly 300 miles above the earth. Thus we see that high- frequency radio waves may travel over short distances in a direct line from the transmitter to the receiver, or they can be radiated upward into the ionosphere to be bent downward in an indirect ray, returning to earth at considerable distance from the transmitter. The wave reaching a receiver via the ionosphere route is termed a sky wave. The wave reaching a receiver by traveling in a direct line from the transmitting antenna to the receiving antenna is commonly called a ground wave. The amount of bending at the ionosphere www.americanradiohistory.com HANDBOOK TRANSMITTING ANTENNA Ground Wave Di DIRECT WAVES e e GROUND- REFLECTED WAVES D2 D3 RECEIVING ANTENNA AT DIFFERENT HEIGHTS Figure 14 INTERFERENCE WITH HEIGHT When the source of a horizontally -polarized space -wave signal is above the horizon, the received signal at a distant location will go through a cyclic variation as the antenna height is progressively raised. This is due to the difference in total path length between the direct wove and the ground-reflected wave, and to the fact that this path length difference changes with antenna height. When the path length difference is such that the two waves arrive at the receiving antenna with a phase difference of 3600 or some multiple of 3600, the two waves will appear WAVE to be in phase as for as the antenna is concerned and maximum signal will be obtained. On the other hand, when the antenna height is such that the path length difference for the two waves causes the waves to arrive with a phase difference of an odd multiple of 1800 the two waves will substantially cancel, and a null will be obtained at that antenna height. The difference between DI and D2 plus D3 is the path- length difference. Note also that there is an additional 1800 phase shift in the ground-reflected wave at the point where it is reflected from the ground. It is this latter phase shift which causes the space-wave field intensity of a horizontally polarized wave to be zero with the receiving antenna at ground level. is in miles and the antenna height N is in feet. This equation must be applied separately to the transmitting and receiving antennas and the results added. However, refraction and diffraction of the signal around the spherical earth cause a smaller reduction in field strength than would occur in the absence of such bending, so that the average radio horizon is somewhat beyond the geometrical horizon. The equation d = 1.4 N,/ f is sometimes used for determining the radio horizon. d Tropospheric Propagation Propagation by signal bending in the lower atmosphere, called tropospheric propagation, can result in the reception of signals over a much greater distance than would be the case if the lower atmosphere were homogeneous. In a homogeneous or well -mixed lower atmosphere, called a normal or standard atmosphere, there is a gradual and uniform decrease in index of refraction with height. This effect is due to Communication 411 the combined effects of a decrease in temperature, pressure, and water -vapor content with height. This gradual decrease in refractive index with height causes waves radiated at very low angles with respect to the horizontal to be bent downward slightly in a curved path. The result of this effect is that such waves will be propagated beyond the true or geometrical horizon. In a so- called standard atmosphere the effect of the curved path is the same as though the radius of the earth were increased by approximately one third. This condition extends the horizon by approximately 30 per cent for normal propagation, and the extendedhorizon is known as the radio path horizon, mentioned before. Conditions Leading to Tropospheric Stratification When the temperature, pressure, or water-vapor content of the atmos- phere does not change smoothly with rising altitude, the discontinuity or stratification will result in the reflection or refraction of incident v -h -f signals. Ordinarily this condition is more prevalent at night and in the summer. In certain areas, such as along the west coast of North America, it is frequent enough to be considered normal. Signal strength decreases slowly with distance and, if the favorable condition in the lower atmosphere covers sufficient area, the range is limited only by the transmitter power, antenna gain, receiver sensitivity, and signal -tonoise ratio. There is no skip distance. Usually, transmission due to this condition is accompanied by slow fading, although fading can be violent at a point where direct waves of about the same strength are also received. Bending in the troposphere, which refers to the region from the earth's surface up to about 10 kilometers, is more likely to occur on days when there are stratus clouds than on clear, cool days with a deep blue sky. The temperature or humidity discontinuities may be broken up by vertical convection currents over land in the daytime but are more likely to continue during the day over water. This condition is in some degree predictable from weather information several days in advance. It does not depend on the sunspot cycle. Like direct communication, best results require similar antenna polarization or orientation at both the transmitting and receiving ends, whereas in transmission via reflection in the ionosphere (that part of the atmosphere between about 50 and 500 kilometers high) it makes little difference whether antennas are similarly polarized. Duct Formation www.americanradiohistory.com When bending conditions are particularly favorable they 41 THE Radiation, Propayation and Lines 0 RADIO bination of the two. The three waves which may combine to make up the ground wave are illustrated in figure 13. @DIRECT WAVE ©GROUND -REFLECTED l--- ' - - - WAVE @SURFACE WAVE Figure - 13 GROUND -WAVE SIGNAL PROPAGATION The illustration above shows the three com- ponents of the ground wave: (A), the surface wave; (B), the direct wave; and (C), the ground-reflected wove. The direct wave and the ground -reflected receiving antenna wove combine at the up the space to make wive. may take place as a result of the ground wave, or as a result of the sky wave or ionospheric wave. The term ground wave actually includes several different types of waves which usually are called: (1) the surface wave, (2) the direct wave, and (3) the ground -reflected wave. The latter two waves combine at the receiving antenna to form the resultant wave or the space wave. The distinguishing characteristic of the components of the ground wave is that all travel along or over the surface of the earth, so that they are affected by the conductivity and terrain of the earth's surface. The Ground Wave Intense bombardment of the upper regions of the atmosphere by radiations from the sun results in the formation of ionized layers. These ionized layers, which form the ionosphere, have the capability of reflecting or refracting radio waves which impinge upon them. A radio wave which has been propagated as a result of one or more reflections from the ionosphere is known as an ionospheric wave or a sky wave. Such waves make possible long distance radio communication. Propagation of radio signals by ionospheric waves is discussed in detail in Sec- The Ionospheric Wove or Sky Wave tion 21 -8. 21 -7 Ground -Wave Communication As stated in the preceding paragraph, the term ground wave applies both to the surface wave and to the space wave (the resultant wave from the combination of the direct wave and the ground -reflected wave) or to a com- The Surface Wove The surface wave is that wave which we normally receive from a standard broadcast station. It travels directly along the ground and terminates on the earth's surface. Since the earth is a relatively poor conductor, the surface wave is attenuated quite rapidly. The surface wave is attenuated less rapidly as it passes over sea water, and the attenuation decreases for a specific distance as the frequency is decreased. The rate of attenuation with distance becomes so large as the frequency is increased above about 3 Mc. that the surface wave becomes of little value for communication. The resultant wave or space wave is illustrated in figure 13 by the combination of (B) and (C). It is this wave path, which consists of the combination of the direct wave and the ground-reflected wave at the receiving antenna, which is the normal path of signal propagation for line -ofsight or near line -of -sight communication or FM and TV reception on frequencies above about 40 Mc. Below line-of -sight over plane earth or water, when the signal source is effectively at the horizon, the ground-reflected wave does not exist, so that the direct wave is the only component which goes to make up the space wave. But when both the signal source and the receiving antenna are elevated with respect to the intervening terrain, the ground-reflected wave is present and adds vectorially to the direct wave at the receiving antenna. The vectorial addition of the two waves, which travel over different path lengths (since one of the waves has been reflected from the ground) results in an interference pattern. The interference between the two waves brings about a cyclic variation in signal strength as the receiving antenna is raised above the ground. This effect is illustrated in figure 14. From this figure it can be seen that best space wave reception of a v -h -f signal often will be obtained with the receiving antenna quite close to the ground. This subject, along with other aspects of v -h -f signal propagation and reception, are discussed in considerable detail in a book on fringe -area TV reception. The distance from an elevated point to the geometrical horizon is gitiren by the approximate equation: d = 1.221 where'the distance The Space Wave "Better TV Reception," by W. W. Smith and R. L. Dowley, published by Editors and Engineers, Ltd., Summer land, Calif. www.americanradiohistory.com HANDBOOK Figure Antenna Bandwidth 409 11 COMPARATIVE VERTICAL RADIATION PATTERNS Showing the vertical radiation patterns of a horizontal single section flat -top beam (A), an array of two stacked horizontal - half -wave elements half of a "Lazy H "-(8), and a horizontal dipole (C). In each case the top of the antenna system is 0.75 wavelength above ground, as shown to the left of in -phase the curves. angle radiation at the expense of the useless high -angle radiation with these simple arrays as contrasted to the dipole is quite marked. Figure 12 compares the patterns of a 3 element beam and a dipole radiator at a height of 0.75 wavelength. It will be noticed that although there is more energy in the lobe of the beam as compared to the dipole, the axis of the beam is at the same angle above the horizontal. Thus, although more radiated energy is provided by the beam at low angles, the average angle of radiation of the beam is no lower than the average angle of radiation of the dipole. 21 -5 21 -6 Propagation of Radio Waves The preceding sections have discussed the manner in which an electromagnetic -wave or radio -wave field may be set up by a radiating system. However, for this field to be useful for communication it must be propagated to some distant point where it may be received, or where it may be reflected so that it may be received at some other point. Radio waves may be propagated to a remote point by either or both of two general methods. Propagation Bandwidth The bandwidth of an antenna or an antenna array is a function primarily of the radiation resistance and of the shape of the conductors which make up the antenna system. For arrays of essentially similar construction the bandwidth (or the deviation in frequency which the system can handle without mismatch) is increased with increasing radiation resistance, and the bandwidth is increased with the use of conductors of larger diameter (smaller ratio of length to diameter). This is to say that if an array of any type is constructed of large diameter tubing or spaced wires, its bandwidth will be greater than that of a similar array constructed of single wires. The radiation resistance of antenna arrays of the types mentioned in the previous paragraphs may be increased through the use of wider spacing between elements. With increased radiation resistance in such arrays the radiation efficiency increases since the ohmic losses within the conductors become a smaller percentage of the radiation resistance, and the bandwidth is increased proportionately. \ A- DIPOLE B-3- ELEMENT PARASITIC 0 1.5 2.0 2.5 3.0 GAIN IN FIELD STRENGTH Figure 3.3 12 VERTICAL RADIATION PATTERNS Showing vertical radiation patterns of a horizontal dipole (A) and a horizontal 3- element parasitic array (8) at a height above ground of 0.75 wavelength. Note that the axis of the main radiation lobes are at the some angle above the horizontal. Note also the suppression of high angle radiation by the parasitic www.americanradiohistory.com array. 408 RADIO THE Radiation, Propagation and Lines Figure 9 VERTICAL RADIATION PATTERNS Showing the vertical radiation patterns for half -wave antennas (or colinear half -wave or extended half-wave antennas) at different heights above average ground and perfect ground. Note that such antennas one -quarter wave above ground concentrate most radiation at the very high angles which are useful for communication only on the lower frequency bands. Antennas one-half wave above ground are not shown, but the elevation pattern shows one lobe on each side at an POWER OUTPUT dipole could be increased by raising the antenna higher above the ground. This is true to an extent in the case of the horizontal dipole; the low -angle radiation does increase slowly after a height of 0.6 wavelength is reached but at the expense of greatly increased high angle radiation and the formation of a number of nulls in the elevation pattern. No signal can be transmitted or received at the elevation angles where these nulls have been formed. Tests have shown that a center height of 0.6 wavelength for a vertical dipole (0.35 wavelength to the bottom end) is about optimum for this type of array. Figure 9 shows the effect of placing a horizontal dipole at various heights above ground. It is easily seen by reference to figure 9 (and figure 10 which shows the radiation from a dipole at ja wave height) that a large percentage of the total radiation from the dipole is being radiated at relatively high angles which are useless for communication on the 14 -Mc. and 28 -Mc. bands. Thus we see that in order to obtain a worthwhile increase in the ratio of low angle radiation to high -angle radiation it is necessary to place the antenna high above ground, and in addition it is necessary to use angle of 30. above horizontal. additional means for suppressing high -angle radiation. High -angle radiation can be suppressed, and this radiation can be added to that going out at low angles, only through the use of some sort of directive antenna system. There are three general types of antenna arrays composed of dipole elements commonly used which concentrate radiation at the lower more effective angles for high -frequency communication. These types are: (1) The close spaced out -of -phase system as exemplified by the "flat -top" beam or a8JK array. Such configurations are classified as end fire arrays. (2) The wide - spaced in -phase arrays, as exemplified by the "Lazy H" antenna. These configurations are classified as broadside arrays. (3) The close- spaced parasitic systems, as exemplified by the three element rotary beam. A comparison between the radiation from a dipole, a "flat -top beam" and a pair of dipoles stacked one above the other (half of a "lazy H "), in each case with the top of the antenna at a height of Sa wavelength is shown in figure 11. The improvement in the amplitude of lowSuppression of High -angle Radiation Figure 10 VERTICAL RADIATION PATTERNS Showing vertical -plane radiation patterns of a horizontal single section flat -top beam with one- eighth wave spacing (solid curves) and a horizontal halfantenna (dashed curves) when both are 0.5 wavelength (A) and 0.75 wavelength (B) a- wav .5 1.0 1.5 2.0 Q.0 3.0 .0 1.0 1.5 2.0 2.0 3.0 GAIN IN FIELD STRENGTH www.americanradiohistory.com bove ground. Antenna HANDBOOK Directivity 407 radiated at other elevation angles is lost and performs no useful function. gy The optimum angle of radiation for propagation of signals between two points is dependent upon a number of variables. Among these significant variables are: (1) height of the ionosphere layer which is providing the reflection, (2) distance between the two stations, (3) number of hops for propagation between the two stations. For communication on the 14 -Mc. band it is often possible for different modes of propagation to provide signals between two points. This means, of course, that more than one angle of radiation can be used. If no elevation directivity is being used under this condition of propagation, selective fading will take place because of interference between the waves arriving over the different paths. On the 28 -Mc. band it is by far the most common condition that only one mode of propagation will be possible between two points at any one time. This explains, of course, the reason why rapid fading in general and selective fading in particular are almost absent from signals heard on the 28 -Mc. band (except for fading caused by local effects). Measurements have shown that the angles useful for communication on the 14-Mc. band are from 3° to about 30 °; angles above about 15° being useful only for local work. On the 28 -Mc. band measurements have shown that the useful angles range from about 3° to 18 °; angles above about 12° being useful only for local (less than 3000 miles) work. These figures assume normal propagation by virtue of the 1:2 layer. Optimum Angle of Radiation .2 .4 .3 .2 .1 0 30 22 26 24 22 20 M N 14 12 IS WAVE ANGLE IN DEGREES 2 0 Figure 8 DIRECTIONAL CHARACTERISTICS OF HORIZONTAL AND VERTICAL DOUBLETS ELEVATED 0.6 WAVELENGTH AND ABOVE TWO TYPES OF VERTICALPLANE GROUND H, represents a horizontal doublet over typical farmland. H2 over salt water. VI is a vertical pattern of radiation from o vertical doublet over typical farmland, V2 over salt water. A salt water ground is the closest approach to an extensive ideally perfect ground that will be met in actual practice. great -circle path, or within 2 or 3 degrees of that path under all normal propagation conditions. However, under turbulent ionosphere conditions, or when unusual propagation conditions exist, the deviation from the great -circle path for greatest signal intensity may be as great as 90 °. Making the array rotatable overcomes these difficulties, but arrays having extremely high horizontal directivity become too cumbersome to be rotated, except perhaps when designed for operation on frequencies above 50 Mc. Vertical directivity is of the great est importance in obtaining satisfactory communication above 14 Mc. whether or not horizontal directivity is used. This is true simply because only the energy radiated between certain definite elevation angles is useful for communication. EnerVertical Directivity Angle of Radiation of Typical Antennos and Arrays It now becomes of interest to determine the smount of radiation available at these useful lower angles of radiation from commonly used an- tennas and antenna arrays. Figure 8 shows relative output voltage plotted against elevation angle (wave angle) in degrees above the horizontal, for horizontal and vertical doublets elevated 0.6 wavelength above two types of ground. It is obvious by inspection of the curves that a horizontal dipole mounted at this height above ground (20 feet on the 28 -Mc. band) is radiating only a small amount of energy at angles useful for communication on the 28 -Mc. band. Most of the energy is being radiated uselessly upward. The vertical antenna above a good reflecting surface appears much better in this respect -and this fact has been proven many times by actual installations. It might immediately be thought that the amount of radiation from a horizontal or vertical www.americanradiohistory.com 406 is resistance of the wire, ground resistance (in the case of a Marconi), corona discharge, and insulator losses. The approximate effective radiation efficiency (expressed as a decimal) is equal to: Nr = Ra /(Ra+ RL) where R. is equal to the radiation resistance and RL is equal to the effective loss resistance of the antenna. The loss resistance will be of the order of 0.25 ohm for large- diameter tubing conductors such as are most commonly used in multi- element parasitic arrays, and will be of the order of 0.5 to 2.0 ohms for arrays of normal construction using copper wire. When the radiation resistance of an antenna or array is very low, the current at a voltage node will be quite high for a given power. Likewise, the voltage at a current node will be very high. Even with a heavy conductor and excellent insulation, the losses due to the high voltage and current will be appreciable if the radiation resistance is sufficiently low. Usually, it is not considered desirable to use an antenna or array with a radiation resistance of less than approximately 5 ohms unless there is sufficient directivity, compactness, or other advantage to offset the losses resulting from the low radiation resistance. The radiation resistance of a Marconi antenna, especially, should be kept as high as possible. This will reduce the antenna current for a given power, thus minimizing loss resulting from the series resistance offered by the earth connection. The radiation resistance can be kept high by making the Marconi radiator somewhat longer than a quarter wave, and shortening it by series capacitance to an electrical quarter wave. This reduces the current flowing in the earth connection. It also should be removed from ground as much as possible (vertical being ideal). Methods of minimizing the resistance of the earth connection will be found in the discussion of the Marconi antenna. Ground Resistance 21 -4 THE Radiation, Propagation and Lines Antenna Directivity All practical antennas radiate better in some directions than others. This characteristic is called directivity. The more directive an antenna is, the more it concentrates the radiation in a certain direction, or directions. The more the radiation is concentrated in a certain direction, the greater will be the field strength produced in that direction for a given amount of total radiated power. Thus the use of a directional antenna or array produces the same result in the favored direction as an increase in the power of the transmitter. The increase in radiated power in a certain RADIO direction with respect to an antenna in free space as a result of inherent directivity is called the free space directivity power gain or just space directivity gain of the antenna (referred to a hypothetical isotropic radiator which is assumed to radiate equally well in all directions). Because the fictitious isotropic radiator is a purely academic antenna, not physically realizable, it is common practice to use as a reference antenna the simplest ungrounded resonant radiator, the half -wave Hertz, or resonant doublet. As a half-wave doublet has a space directivity gain of 2.15 db over an isotropic radiator, the use of a resonant dipole as the comparison antenna reduces the gain figure of an array by 2.15 db. However, it should be understood that power gain can be expressed with regard to any antenna, just so long as it is specified. As a matter of interest, the directivity of an infinitesimal dipole provides a free space directivity power gain of 1.5 (or 1.76 db) over an isotropic radiator. This means that in the direction of maximum radiation the infinitesimal dipole will produce the same field of strength as an isotropic radiator which is radiating 1.5 times as much total power. A half -wave resonant doublet, because of its different current distribution and significant length, exhibits slightly more free space power gain as a result of directivity than does the infinitesimal dipole, for reasons which will be explained in a later section. The space directivity power gain of a half -wave resonant doublet is 1.63 (or 2.15 db) referred to an isotropic radiator. choosing and orienting an antenna system, the radiation patterns of the various common types of antennas should be given careful consideration. The directional characteristics are of still greater importance when a directive antenna array is used. Horizontal directivity is always desirable on any frequency for point -to -point work. However, it is not always attainable with reasonable antenna dimensions on the lower frequencies. Further, when it is attainable, as on the frequencies above perhaps 7 Mc., with reasonable antenna dimensions, operating convenience is greatly furthered if the maximum lobe of the horizontal directivity is controllable. It is for this reason that rotatable antenna arrays have come into such common usage. Considerable horizontal directivity can be used to advantage when: (1) only point -topoint work is necessary, (2) several arrays are available so that directivity may be changed by selecting or reversing antennas, (3) a single rotatable array is in use. Signals follow the Horizontal Directivity When www.americanradiohistory.com HANDBOOK Antenna Impedance HEIGHT IN WAVELENGTHS OF CENTER OF VERTICAL HALF -WAVE ANTENNA ABOVE PERFECT GROUND 25 .3 .4 .5 .5 .7 .75 V 0 .1 w-_ 571 .2 -MON OH7AL .3 .4 .S 41 .7 . .5 to HEIGHT IN WAVELENGTHS OF HORIZONTAL HALF WAVE ANTENNA ABOVE PERFECT GROUND Figure 7 EFFECT OF HEIGHT ON THE RADIATION RESISTANCE OF A DIPOLE SUSPENDED ABOVE PERFECT GROUND the radiation resistance to approximately 100 ohms. When a horizontal half -wave antenna is used, the radiation resistance (and, of course, the amount of energy radiated for a given antenna current) depends on the height of the antenna above ground, since the height determines the phase and amplitude of the wave reflected from the ground back to the antenna. Thus the resultant current in the antenna for a given power is a function of antenna height. linear radiator is series fed the center, the resistive and reactive components of the driving point impedance are dependent upon both the length and diameter of the radiator in wavelengths. The manner in which the resistive component varies with the physical dimensions of the radiator is illustrated in figure 5. The manner in which the reactive component varies is illustrated in figure 6. Several interesting things will be noted with respect to these curves. The reactive component disappears when the overall physical length is slightly less than any number of half waves long, the differential increasing with conductor diameter. For overall lengths in the vicinity of an odd number of half wavelengths, the center feed point looks to the generator or transmission line like a series -resonant lumped circuit, while for overall lengths in the vicinity of an even number of half wavelengths, it looks like a parallel- resonant or anti- resonant lumped circuit. Both the feed point resistance Center -fed When a Feed Point Impedance at 405 and the feed point reactance change more slowly with overall radiator length (or with frequency with a fixed length) as the conductor diameter is increased, indicating that the effective "Q" is lowered as the diameter is in- creased. However, in view of the fact that the damping resistance is nearly all "radiation resistance" rather than loss resistance, the lower Q does not represent lower efficiency. Therefore, the lower Q is desirable, because it permits use of the radiator over a wider frequency range without resorting to means for eliminating the reactive component. Thus, the use of a large diameter conductor makes the overall system less frequency sensitive. If the diameter is made sufficiently large in terms of wavelengths, the Q will be low enough to qualify the radiator as a "broad- band" antenna. The curves of figure 7 indicate the theoretical center -point radiation resistance of a half wave antenna for various heights above perfect ground. These values are of importance in matching untuned radio -frequency feeders to the antenna, in order to obtain a good impedance match and an absence of standing waves on the feeders. Ground Losses Above average ground, the actual radiation resistance of a dipole will vary from the exact value of figure 7 since the latter assumes a hypothetical, perfect ground having no loss and perfect reflection. Fortunately, the curves for the radiation resistance over most types of earth will correspond rather closely with those of the chart, except that the radiation resistance for a horizontal dipole does not fall off as rapidly as is indicated for heights below an eighth wavelength. However, with the antenna so close to the ground and the soil in a strong field, much of the radiation resistance is actually represented by ground loss; this means that a good portion of the antenna power is being dissipated in the earth, which, unlike the hypothetical perfect ground, has resistance. In this case, an appreciable portion of the radiation resistance actually is loss resistance. The type of soil also has an effect upon the radiation pattern, especially in the vertical plane, as will be seen later. The radiation resistance of an antenna generally increases with length, although this increase varies up and down about a constantly increasing average. The peaks and dips are caused by the reactance of the antenna, when its length does not allow it to resonate at the operating frequency. Antennas have a certain loss resistance as well as a radiation resistance. The loss resistance defines the power lost in the antenna due to ohmAntenna Efficiency www.americanradiohistory.com 404 THE Radiation, Propagation and Lines 10000 +9000 9000 +5000 8000 +4000 7000 +3000 6000 +2000 + 1000 5000. 4000 DIAMETER= rka000, 3000 2000 i =r' 2000 l DIAMETER1000 0 O.15Á 05A 1.05 1.5A 2.05 3000 4000 2.55 OVERALL LENGTH OF RADIATOR Figure 5 FEED POINT RESISTANCE OF A CENTER DRIVEN RADIATOR AS A FUNCTION OF PHYSICAL LENGTH IN TERMS OF FREE SPACE WAVELENGTH When the antenna is resonant, and it always should be for best results, the impedance at the center is substantially resistive, and is termed the radiation resistance. Radiation resistance is a fictitious term; it is that value of resistance (referred to the current loop) which would dissipate the same amount of power as being radiated by the antenna, when fed with the current flowing at the current loop. The radiation resistance depends on the antenna length and its proximity to nearby objects which either absorb or re- radiate power, such as the ground, other wires, etc. The Marconi Antenna RADIO Before going too far with the discussion of radiation resistance, an explanation of the Marconi (grounded quarter wave) antenna is in order. The Marconi antenna is a special type of Hertz antenna in which the earth acts as the "other half" of the dipole. In other words, the current flows into the earth instead of into a similar quarter -wave section. Thus, the current loop of a Marconi antenna is at the base rather than in the center. In either case it is a quarter wavelength from the end. A half -wave dipole far from ground and other reflecting objects has a radiation resistance at the center of about 73 ohms. A Marconi an- 5000 0 155 055 OA 1.25 2.0 OVERALL LENGTH OF RADIATOR I 2.55 Figure 6 REACTIVE COMPONENT OF THE FEED A CENTER IMPEDANCE OF POINT DRIVEN RADIATOR AS A FUNCTION OF PHYSICAL LENGTH IN TERMS OF FREE SPACE WAVELENGTH tenna is simply one -half of a dipole. For that reason, the radiation resistance is roughly half the 73-ohm impedance of the dipole or 36.5 ohms. The radiation resistance of a Marconi antenna such as a mobile whip will be lowered by the proximity of the automobile body. Because the power throughout the antenna is the same, the impedance of a resonant antenna at any point along its length merely expresses the ratio between voltage and current at that point. Thus, the lowest impedance occurs where the current is highest, namely, at the center of a dipole, or a quarter wave from the end of a Antenna Impedance Marconi. The impedance rises uniformly toward each end, where it is about 2000 ohms for a dipole remote from ground, and about twice as high for a vertical Marconi. If a vertical half-wave antenna is set up so that its lower end is at the ground level, the effect of the ground reflection is to increase www.americanradiohistory.com HANDBOOK Radiation Resistance A harmonic operated antenna is somewhat longer than the corresponding integral number of dipoles, and for this reason, the dipole length formula cannot be used simply by multiplying by the corresponding harmonic. The intermediate half wave sections do not have end effects. Also, the current distribution is disturbed by the fact that power can reach some of the half wave sections only by flowing through other sections, the latter then acting not only as radiators, but also as transmission lines. For the latter reason, the resonant length will be dependent to an extent upon the method of feed, as there will be less attenuation of the current along the antenna if it is fed at or near the center than if fed towards or at one end. Thus, the antenna would have to be somewhat longer if fed near one end than if fed near the center. The difference would be small, however, unless the antenna were many wavelengths long. The length of a center fed harmonically operated doublet may be found from the formula: L (K -.05) x 492 Freq. in Mc. Under conditions of severe current attenuation, it is possible for some of the nodes, or loops, actually to be slightly greater than a physical half wavelength apart. Practice has shown that the most practical method of resonating a harmonically operated antenna accurately is by cut and try, or by using a feed system in which both the feed line and antenna are resonated at the station end as an integral system. A dipole or half-wave antenna is said to operate on its fundamental or first harmonic. A full wave antenna, 1 wavelength long, operates on its second harmonic. An antenna with five half- wavelengths on it would be operating on its fifth harmonic. Observe that the fifth harmonic antenna is 2tfs wavelengths long, not wavelengths. Antenna Resonance 000 Di o.sa rv=. Figure 4 EFFECT OF SERIES INDUCTANCE AND CAPACITANCE ON THE LENGTH OF A HALF -WAVE RADIATOR The top antenna has been electrically lengthened by placing o coil in series with the center. In other words, an antenna with a lumped inductance in its center can be mode shorter for a given frequency than a plain wire radiator. The bottom antenna has been capacitively shortened electrically. In other words, on antenna with o capacitor in series with it must be mode longer for o given frequency since its effective electrical length os compared to plain wire is shorter. i where K = number of waves on antenna L = length in feet 5 403 Most types of antennas operate most efficiently when tuned or resonated to the frequency of operation. This consideration of course does not apply to the rhombic antenna and to the parasitic elements of arrays employing parasitically excited elements. However, in practically every other case it will be found that increased efficiency results when the entire antenna system is resonant, whether it be a simple dipole or an elaborate array. The radiation efficiency of a resonant wire is many times that of a wire which is not resonant. If an antenna is slightly too long, it can be resonated by series insertion of a variable capacitor at a high current point. If it is slightly too short, it can be resonated by means of a variable inductance. These two methods, illustrated schematically in figure 4, are generally employed when part of the antenna is brought into the operating room. With an antenna array, or an antenna fed by means of a transmission line, it is more common to cut the elements to exact resonant length by "cut and try" procedure. Exact antenna resonance is more important when the antenna system has low radiation resistance; an antenna with low radiation resistance has higher Q (tunes sharper) than an antenna with high radiation resistance. The higher Q does not indicate greater efficiency; it simply indicates a sharper resonance curve. Radiation Resistance 21 -3 and Feed -Point Impedance In many ways, a half-wave antenna is like a tuned tank circuit. The main difference lies in the fact that the elements of inductance, capacitance, and resistance are lumped in the tank circuit, and are distributed throughout the length of an antenna. The center of a half -wave radiator is effectively at ground potential as far as r-f voltage is concerned, although the current is highest at that point. www.americanradiohistory.com 402 RADIO THE Radiation, Propagation and Lines distance in meters between adjacent peaks or adjacent troughs of a wave train. As a radio wave travels 300,000,000 meters a second (speed of light), a frequency of cycle per second corresponds to a wavelength 1 of 300,000,000 meters. So, if the frequency is multiplied by a million, the wavelength must be divided by a million, in order to maintain their correct ratio. A frequency of 1,000,000 cycles per second (1,000 kc.) equals a wavelength of 300 meters. Multiplying frequency by 10 and dividing wavelength by 10, we find: a frequency of 10,000 kc. equals a wavelength of 30 meters. Multiplying and dividing by 10 again, we get: a frequency of 100,000 kc. equals 3 meters wavelength. Therefore, to change wavelength to frequency (in kilocycles), simply divide 300,000 by the wavelength in meters (À). 300,000 Fkc = À À - 300,000 Fkc Now that we have a simple conversion formula for converting wavelength to frequency and vice versa, we can combine it with our wavelength versus antenna length formula, and we have the following: Length of a half -wave radiator made from wire (no. 14 to no. 10): 3.5 -11c. to 30 -Mc. bands 468 Length in feet = Freq. in Mc. 50 -Mc. band Length in feet Length in inches - 40 .40 130 200 100 RATIO OF 300 X00 400,000 400 shortening can be determined with the aid of the chart of figure 3. In this chart the amount of additional shortening over the values given in the previous paragraph is plotted against the ratio of the length to the diameter of the half -wave radiator. The length of a wave in free space is somewhat longer than the length of an antenna for we same frequency. The actual free -space half- wavelength is given by the following expressions: 460 Freq. in Mc. Half- wavelength = 5600 Freq. in Mc. Half- wavelength 492 Freq. in Mc. 5905 w 5500 Mc. half -wave radiator is constructed from tubing or rod whose diameter is an appreciable fraction of the length of the radiator, the resonant length of a half -wave antenna will be shortened. The amount of Length -to- Diameter Ratio When a Mc. in feet in inches wire in space can resonate at more than one frequency. The lowest frequency at which it resonates is called its fundamental frequency, and at that frequency it is approximately a half wavelength long. A wire can have two, three, four, five, or more standing waves on it, and thus it resonates at approximately the integral harmonics of its fundamental frequency. However, the higher harmonics are not exactly integral multiples of the lowest resonant frequency as a result of end effects. Harmonic Resonance 144 -Mc. band Freq. in .00 Figure 3 CHART SHOWING SHORTENING OF A ELEMENT IN TERMS OF RESONANT RATIO OF LENGTH TO DIAMETER The use of this chart is based on the basic formula where radiator length in feet is equal to 468 /frequency in Mc. This formula applies to frequencies below perhaps 30 Mc. when the radiator is made from wire. On higher frequencies, or on 14 and 28 Mc. when the radiator is made of large- diameter tubing, the radiator is shortened from the value obtained with the above formula by on amount determined by the ratio of length to diameter of the radiator. The amount of this shortening is obtainable from the chart shown above. Freq. in Length in inches = :000 LENGTH TO DIAMETER A www.americanradiohistory.com HANDBOOK Antenna Characteristics 401 Figure 2 ANTENNA POLARIZATION The polarization (electric field) of the radiation from a resonant dipole such as shown at (A) above is parallel to the length of the radiator. In the case of o resonant slot cut in a sheet of metal and used as a radiator, the polarization (of the electric field) is perpendicular to the length of the slot. In both cases, however, the polarization of the radiated field is parallel to the potential gradient of the radiator; in the case of the dipole the electric lines of force are from end to end, while in the case of the slot the field is across the sides of the slot. The metallic sheet containing the slot may be formed into a cylinder to make up the radiator shown at (C). With this type of radiator the radiated field will be horizontally polarized even though the radiator is mounted vertically. is a graph showing the relative radiated field intensity against azimuth angle for horizontal directivity and field intensity against elevation angle for vertical directivity. The bandwidth of an antenna is a measure of its ability to operate within specified limits over a range of frequencies. Bandwidth can be expressed either "operating frequency plus or -minus a specified per cent of operating frequency" or "operating frequency plus -or -minus a specified number of megacycles" for a certain standing- wave -ratio limit on the transmission line feeding the antenna system. The effective power gain or directive gain of an antenna is the ratio between the power required in the specified antenna and the power required in a reference antenna (usually a halfwave dipole) to attain the same field strength in the favored direction of the antenna under measurement. Directive gain may be expressed either as an actual power ratio, or as is more common, the power ratio may be expressed in decibels. Physical Length of a Half -Wave cross section of the conductor which makes up Antenna the antenna is kept very small with respect to the antenna length, an electrical half wave is a fixed percentage shorter than a physical halfwavelength. This percentage is approximately 5 per cent. Therefore, most linear half-wave antennas are close to 95 per cent of a half wavelength long physically. Thus, a half-wave antenna resonant at exactly 80 meters would be one -half of 0.95 times 80 meters in length. Another way of saying the same thing is that a If the ELECTRIC FIELD (POLARIZATION) VERTICAL O ELECTRIC FIELD (POLARIZATION) HORIZONTAL .or FEEDERS CONNECT TO POINTS Aas NSIDC CYLINDER wire resonates at a wavelength of about 2.1 times its length in meters. If the diameter of the conductor begins to be an appreciable fraction of a wavelength, as when tubing is used as a v -h -f radiator, the factor becomes slightly less than 0.95. For the use of wire and not tubing on frequencies below 30 Mc., however, the figure of 0.95 may be taken as accurate. This assumes a radiator removed from surrounding objects, and with no bends. Simple conversion into feet can be obtained by using the factor 1.56. To find the physical length of a half -wave 80 -meter antenna, we multiply 80 times 1.56, and get 124.8 feet for the length of the radiator. It is more common to use frequency than wavelength when indicating a specific spot in the radio spectrum. For this reason, the relationship between wavelength and frequency must be kept in mind. As the velocity of radio waves through space is constant at the speed of light, it will be seen that the more waves that pass a point per second(higher frequency), the closer together the peaks of those waves must be (shorter wavelength). Therefore, the higher the frequency, the lower will be the wavelength. A radio wave in space can be compared to a wave in water. The wave, in either case, has peaks and troughs. One peak and one trough constitute a full wave, or one wavelength. Frequency describes the number of wave cycles or peaks passing a point per second. Wavelength describes the distance the wave travels through space during one cycle or oscillation of the antenna current; it is the www.americanradiohistory.com THE Radiation, Propagation and Lines 400 ` VOLTAGE . ,---t--_ me/m..1.5,-k\ C[NTCII ' ` `{ . 1...-14ALW-WAVE ANTENNA f i . SHOWING NOW STANDING WAVES CRUST ON A HORIZONTAL ANTENNA. C. VOLTAGE CURRENT IS MAXIMUM AT CENTRA. VOLTAGE IS MAXIMUM AT Figure STANDING WAVES ON A RESONANT ANTENNA 1 transmission lines, both from single -wire lines and from lines comprised of more than one wire. In addition, radiation can be made to take place in a very efficient manner from electromagnetic horns, from plastic lenses or from electromagnetic lenses made up of spaced conducting planes, from slots cut in a piece of metal, from dielectric wires, or from the open end of a wave guide. Directivity of Radiation The radiation from any phys- ically practicable radiating system is directive to a certain degree. The degree of directivity can be enhanced or altered when desirable through the combination of radiating elements in a prescribed manner, through the use of reflecting planes or curved surfaces, or through the use of such systems as mentioned in the preceding paragraph. The construction of directive antenna arrays is covered in detail in the chapters which follow. Like light waves, radio waves can have a definite polarization. In fact, while light waves ordinarily have to be reflected or passed through a polarizing medium before they have a definite polarization, a radio wave leaving a simple radiator will have a definite polarization, the polarization being indicated by the orientation of the electric -field component of the wave. This, in turn, is determined by the orientation of the radiator itself, as the magnetic -field component is always at right angles to a linear radiator, and the electric -field component is always in the same plane as the radiator. Thus we see that an antenna that is vertical with respect to the earth will transmit a vertically polarized wave, as the electrostatic lines of force will be vertical. Likewise, a simple horizontal antenna will radiate horizontally polarized waves. Polarization RADIO Because the orientation of a simple linear radiator is the same as the polarization of the waves emitted by it, the radiator itself is referred to as being either vertically or horizontally polarized. Thus, we say that a horizontal antenna is horizontally polarized. Figure 2A illustrates the fact that the polarization of the electric field of the radiation from a vertical dipole is vertical. Figure 2B, on the other hand, shows that the polarization of electric -field radiation from a vertical slot radiator is horizontal. This fact has been utilized in certain commercial FM antennas where it is desired to have horizontally polarized radiation but where it is more convenient to use an array of vertically stacked slot arrays. If the metallic sheet is bent into a cylinder with the slot on one side, substantially omnidirectional horizontal coverage is obtained with horizontally -polarized radiation when the cylinder with the slot in one side is oriented vertically. An arrangement of this type is shown in figure 2C. Several such cylinders may be stacked vertically to reduce high -angle radiation and to concentrate the radiated energy at the useful low radiation angles. In any event the polarization of radiation from a radiating system is parallel to the electric field as it is set up inside or in the vicinity of the radiating system. 21 -2 General Character- istics of Antennas antennas have certain general characterIt is the result of differences in these general characteristics which makes one type of antenna system most suitable for one type of application and another type best for a different application. Six All istics to be enumerated. of the more important characteristics are: (1) polarization, (2) radiation resistance, (3) horizontal directivity, (4) vertical directivity, (5) bandwidth, and (6) effective power gain. The polarization of an antenna or radiating system is the direction of the electric field and has been defined in Section 21 -1. The radiation resistance of an antenna system is normally referred to the feed point in an antenna fed at a current loop, or it is referred to a current loop in an antenna system fed at another point. The radiation resistance is that value of resistance which, if inserted in series with the antenna at a current loop, would dissipate the same energy as is actually radiated by the antenna if the antenna current at the feed point were to remain the same. The horizontal and vertical directivity can best be expressed as a directive pattern which www.americanradiohistory.com CHAPTER TWENTY -ONE Radiation, Propagation and Transmission Lines Radio waves are electromagnetic waves similar in nature but much lower in frequency than light waves or heat waves. Such waves represent electric energy traveling through space. Radio waves travel in free space with the velocity of light and can be reflected and refracted much the same as light waves. 21 -1 Radiation from possible change in the electrical constants of a line is that which occurs at the open end of a wire. Therefore, a dipole has a great mismatch at each end, producing a high degree of reflection. We say that the ends of a dipole are terminated in an infinite impedance. A returning wave which has been reflected meets the next incident wave, and the voltage and current at any point along the antenna are the vector sum of the two waves. At the ends of the dipole, the voltages add, while the currents of the two waves cancel, thus producing high voltage and low current at the ends of the dipole or half wave section of wire. In the same manner, it is found that the currents add while the voltages cancel at the center of the dipole. Thus, at the center there is high current but low voltage. Inspection of figure 1 will show that the current in a dipole decreases sinusoidally towards either end, while the voltage similarly increases. The voltages at the two ends of the antenna are 180° out of phase, which means that the polarities are opposite, one being plus while the other is minus at any instant. A curve representing either the voltage or current on a dipole represents a standing wave on the wire. an Antenna Alternating current passing through a conductor creates an alternating electromagnetic field around that conductor. Energy is alternately stored in the field, and then returned to the conductor. As the frequency is raised, more and more of the energy does not return to the conductor, but instead is radiated off into space in the form of electromagnetic waves, called radio waves. Radiation from a wire, or wires, is materially increased whenever there is a sudden change in the electrical constants of the line. These sudden changes produce reflection, which places standing waves on the line. When a wire in space is fed radio frequency energy having a wavelength of approximately 2.1 times the length of the wire in meters, the wire resonates as a half-wave dipole antenna at that wavelength or frequency. The greatest Radiation from Sources other than Antennas 399 www.americanradiohistory.com Radiation can and does take place from sources other than antennas. Undesired radiation can take place from open -wire 3 98 OSCILLATOR BUFFER i BUFFER Sta I +300 V. OUTPUT CONTROL ° RI 100E 2w -120V Figure 19 DIFFERENTIAL KEYING SYSTEM WITH OSCILLATOR SWITCHING DIODE Vi V2 OSCILLATOR V3 DRIVER BUFFER 300 V. n ° V4 12AU7 100 It REVER TUBE 6 22 REV 005 100 E R2 R3 100E 4TE VFO"MOLD -50V 005 100E 330E C11`OS Figure 20 DIFFERENTIAL KEYER EMPLOYED IN "JOHNSON" TRANSMITTERS conducting--and then continue operating until atter V2 and V3 have stopped conducting. Potentiometer R1 adjusts the "hold" time for VFO operation after the key is opened. This may be adjusted to cut off the VFO between marks of keyed characters, thus allowing rapid break -in operation. www.americanradiohistory.com Differential HANDBOOK Keying 6AL5 u BLOCKING DIODES o 397 TO CATHODE CIRCUIT OF KEYED STAGE Ó O O f r u 2 o VI -250 170 Pt V. VACUUM TUBE KEYER U ñ (FIG. Po) 1 ó -Y-CUr-OFF VALUE AMPLIFIER I \ CUT -OFF VALUE OSC. Figure BLOCKING DIODES EMPLOYED TO VARY TIME CONSTANT OF "MAKE" AND "BREAK" CHARACTERISTICS OF VACUUM TUBE KEYER -- DURING DEPRESSED KEY iS THIS TIME 6-TRANSMITTER IS YON THE AIR- DURING THIS TIME Figure 18 17 TIME SEQUENCE OF A DIFFERENTIAL KEYER on a moment before the rest of the stages are energized, and remains on a moment longer than the other stages. The "chirp" or frequency shift associated with abrupt switching of the oscillator is thus removed from the emitted signal. In addition, the differential keyer can apply waveshaping to the amplifier section of the transmitter, eliminating the "click" caused by rapid keying of the latter stages. The ideal keying system would perform as illustrated in figure 17. When the key is closed, the oscillator reaches maximum output almost instantaneously. The following stages reach maximum output in a fashion determined by the waveshaping circuits of the keyer. When the key is released, the output of the amplifier stages starts to decay in a predetermined manner, followed shortly thereafter by cessation of the oscillator. The overall result of these actions is to provide relatively soft "make" and "break" to the keyed signal, meanwhile preventing oscillator frequency shift during the keying sequence. The rates of charge and decay in a typical R -C keying circuit may be varied independently of each other by the blocking diode system of figure 18. Each diode permits the charging current of the timing capacitor to flow through only one of the two variable potentiometers, thus permitting independent adjustment of the "make" and "break" characteristics of the keying system. A practical differential keying system de- veloped by WIICP (Feb., 1956 QST) is shown in figure 19. A 6AL5 switch tube turns the oscillator on before the keying action starts, and holds it on until after the keying sequence is completed. Time constant of the keying cycle is determined by values of C and R. When the key is open, a cut -off bias of about -110 volts is applied to the screen grid circuits of the keyed stages. When the key is closed, the screen grid voltage rises to the normal value at a rate determined by the time constant R -C. Upon opening the key again, the screen voltage returns to cut -off value at the predetermined rate. The potentiometer R1 serves as an output control, varying the minimum internal resistance of the 12BH7 keyer tube, and is a useful device to limit power input during tune up periods. Excitation to the final amplifier stage may be controlled by the screen potentiometer R3 in the second buffer stage. An external bias source of approximately -120 volts at 10 milliamperes is required for operation of the keyer, in addition to the 300-volt screen supply. Blocking voltage may be removed from the oscillator for "zeroing" purposes by closing switch Si, rendering the diode switch in- operative. keying system is shown figure 20, and is widely used in many Johnson transmitters. Grid block keying is used on tubes V2 and V3. A waveshaping filter consisting of R2, R3, and C1 is used in the keying control circuit of V2 and V3. To avoid chirp when the oscillator (V1) is keyed, the keyer tube V4 allows the oscillator to start quickly before V2 and V3 start A second popular in www.americanradiohistory.com -- 396 Transmitter Keying and Control THE RADIO LOW POWER SUFFER 6AG7 'wet +M.V. KEYER UNIT &LOCK /NG 64/0 VOLTAGE VOLTS t TIME OUTPUT TO SCREEN CIS 807 - r O+-+ + VOLTS - TIME 6116 .025 470K,1 W K Io W KEY VP KEY DOWN A -35 340 B -Ito 0 C -no 0 D 375 375 E -275 -273 D+ 4 0-5 Io POINT 10M.S0MA. 3 25K1 IOW 6Ax5- GT 'n81 ce 450V 5Y3 a OAKS v. 12AÚ7 SKIS MTR. AN 4.70,2W 350-0-350 50 MA. TWO -STAGE Figure 16 SCREEN GRID KEYER guished, removing the screen voltage from the tetrode r -f tube. At the same time, rectified grid bias is applied to the screen of the tetrode through the I megohm resistor between screen and key. This voltage effectively cuts off the screen of the tetrode until the key is closed again. The RC circuit in the grid of the 6L6 tube determines the keying characteristic of the tetrode tube. A more elaborate screen grid keyer is shown in figures 15 and 16. This keyer is designed to block -grid key the oscillator or a low powered buffer stage, and to screen key a medium powered tetrode tube such as an 807, 2E26 or 6146. The unit described includes a simple dual voltage power supply for the positive screen voltage of the tetrode, and a negative supply for the keyer stages. A 6K6 is used as the screen keyer, and a 12AU7 is used as a cathode follower and grid block keyer. As in UNIT the W1DX keyer, this keyer turns on the exciter a moment before the tetrode stage is turned on. The tetrode stage goes off an instant before the exciter does. Thus any keying chirp of the oscillator is effectively removed from the keyed signal. By listening in the receiver one can hear the exciter stop operating a fraction of a second after the tetrode stage goes off. In fact, during rapid keying, the exciter may be heard as a steady signal in the receiver, as it has appreciable time lag in the keying circuit. The clipping effect of following stages has a definite hardening effect on this, however. 20 -8 Differential Keying Circuits Excellent waveshaping may be obtained by differential keying system whereby the master oscillator of the transmitter is turned a www.americanradiohistory.com Screen HANDBOOK 807. Keying EXC. 20.7 14 SINGLE -STAGE SCREEN GRID KEYER FOR TETRODE TUBES tetrode is keyed by this method, there is the possibility of a considerable backwave caused by r -f leakage through the grid -plate capacity of the tube. Certain hi -µ triode tubes, such as the 811 -A and the 805, automatically block themselves when the grid return circuit is opened. It is merely necessary to insert a key and associated key click filter in the grid return lead of these tubes. No blocking bias supply is needed. This circuit is shown in figure 12. A more elaborate blocked -grid keying system has been developed by W1DX, and was shown in the February, 1954 issue of QST magazine. This highly recommended circuit is shown in figure 13. Two stages are keyed, Screen Grid Keying The screen circuit of a tetrode tube may be keyed for c -w operation. Unfortunately, when the screen grid of a tetrode tube is brought to zero potential, the tube still delivers considerable output. Thus it is necessary to place a negative blocking voltage on the screen grid to reduce the backwave through the tube. A suitable keyer circuit that will achieve this was developed by W6DTY, and was described in the February, 1953 issue of CQ magazine. This circuit is shown in figure 14. A 6L6 is used as a combined clamper tube and keying tube. When the key is closed, the 6L6 tube has blocking bias applied to its control grid. This bias is obtained from the rectified grid bias of the keyed tube. Screen voltage is applied to the keyed stage through a screen dropping resistor and a VR -105 regulator tube. then the key is open, the 6L6 is no longer cut -off, and conducts heavily. The voltage drop across the dropping resistor caused by the heavy plate current of the 6L6 lowers the voltage on the VR -105 tube until it is extin- I^N Figure 15 TOP VIEW OF SCREEN GRID KEYER SHOWN IN FIGURE 395 preventing any backwave emission. The first keyed stage may be the oscillator, or a low powered buffer. The last keyed stage may be the driver stage to the power amplifier, or the amplifier itself. Since the circuit is so proportioned that the lower powered stage comes on /first and goes off last, any keying chirp in the oscillator is not emitted on the air. Keying lag is applied to the high powered keyed stage only. ETC Figure Grid 16 www.americanradiohistory.com Transmitter Keying 394 THE and Control RADIO HI -MU TRIODE (61 -A ETC.) I H.V -BLOCKING B AS Figure 11 SIMPLE BLOCKED -GRID KEYING SYSTEM The blocking bias must be sufficient to cutoff plate current to the amplifier stage in the presence of the excitation voltage. R¡ is normal bias resistor for the tube. R2 and C1 should be adjusted for correct keying waveform. recommended for general use, as considerable voltage will be developed across the key when it is open. An electronic switch can take the place of the hand key. This will remove the danger of shock. At the same time, the opening and closing characteristics of the electronic switch may easily be altered to suit the particular need at hand. Such an electronic switch is called a vacuum tube keyer. Low internal resistance triode tubes such as the 45, 6A3, or 6AS7 are used in the keyer. These tubes act as a very high resistance when sufficient 807, 6146, ISO LOW POWER BUFFER (5457 ETC.) ETC. LUF RFC 2 SUN 33K 2W 1001t IW +M Figure 12 SELF -BLOCKING KEYING SYSTEM FOR HIGH -MU TRIODE R, and C1 adjusted for correct keying waveform. R, is bias resistor of tube. blocking bias is applied to them, and as a very low resistance when the bias is removed. The desired amount of lag or cushioning effect can be obtained by employing suitable resistance and capacitance values in the grid of the keyer tube(s). Because very little spark is produced at the key, due to the small amount of power in the key circuit, sparking clicks are easily suppressed. One type 45 tube should be used for every 50 ma. of plate current. Type 6B4G or 2A3 tubes may also be used; allow one 6B4G tube for every 80 ma. of plate current. Because of the series resistance of the keyer tubes, the plate voltage at the keyed tube will be from 30 to 60 volts less than the power supply voltage. This voltage appears as cathode bias on the keyed tube, assuming the bias return is made to ground, and should be taken into consideration when providing bias. Some typical cathode circuit vacuum tube keying units are shown in figure 10. V. VR-150 REY -+00 6.3V. TO V. 6J5 Figure 13 BLOCKED-GRID KEYER A separate filament transformer must be used for the 6J5, as its filament is at a potential of -400 volts. TWO -STAGE 20-6 Grid Circuit Keying Grid circuit, or blocked grid keying is another effective method of keying a c -w transmitter. A basic blocked grid keying circuit is shown in figure 11. The time constant of the keying is determined by the RC circuit, which also forms part of the bias circuit of the tube. When the key is closed, operating bias is developed by the flow of grid current through 121. When the key is open, sufficient fixed bias is applied to the tube to block it, preventing the stage from functioning. If an un- neutralized HANDBOOK Cathode TO SOMA SELENIUM RECTIFIER 7. I 6Y6 IM (BREAK) I M(MAKE) Keying 393 RF TUBE CATHODE OF KEYED STAGE I W STANCOR PAB421 45/2A3 2K ,2W 350.0 -350 70 SOMA IM 47014 IW I a°) 4 e00V RFC Q_ooe 2.SMH 1 TY705 MI BO KEY 45/2A3 4704,0.1W 1 "T IoODUF t M.Y. SUPPLY G Figure 10 VACUUM TUBE KEYERS FOR CENTER -TAP KEYING CIRCUITS The type A keyer is suitable for keying stages running up to 1250 volts on the plate. Two 2A3 or 6A3 tubes can safely key 160 milliamperes of cathode current. The simple 6Y6 keyer in figure B Is for keying stages running up to 650 volts on the plate. A single 6Y6 can key 80 milliamperes. Two in parallel may be used for plate currents under 160 ma. If softer keying is desired, the 500 -µofd. mica condenser should be increased to .001 pfd. amplifier. If a low -level stage, which is followed by a series of class C amplifiers, is keyed, serious transients will be generated in the output of the transmitter even though the keyed stage is being turned on and off very smoothly. This condition arises as a result of pulse sharpening, which has been discussed previously. Third, the output from the stage should be completely cut off when the key is up, and the time constant of the rise and decay of the keying wave should be easily controllable. Fourth, it should be possible to make the rise period and the decay period of the keying wave approximately equal. This type of keying envelope is the only one tolerable for commercial work, and is equally desirable for obtaining clean cut and easily readable signals in amateur work. Fifth, it is desirable that the keying circuit be usable without a keying relay, even when a high -power stage is being keyed. Last, for the sake of simplicity and safety, it should be possible to ground the frame of the key, and yet the circuit should be such that placing the fingers across the key will not result in an electrical shock. In other words, the keying circuit should be inherently safe. All these requirements have been met in the keying circuits to be described. 20-5 Cathode Keying The lead from the cathode or center -tap connection of the filament of an r -f amplifier can be opened and closed for a keying circuit. Such a keying system opens the plate voltage circuit and at the same time opens the grid bias return lead. For this reason, the grid circuit is blocked at the same time the plate circuit is opened. This helps to reduce the backwave that might otherwise leak through the keyed stage. The simplest cathode keying circuit is illustrated in figure 9, where a key -click filter is employed, and a hand key is used to break the circuit. This simple keying circuit is not www.americanradiohistory.com 392 THE Transmitter Keying and Control RADIO a wide frequency band as sidebands and are heard as clicks. The cure for transient key clicks is relatively simple, although one would not believe it, judging from the hordes of clicky, "snappy" signals heard on the air. To be capable of transmitting code characters and at the same time not splitting the eardrums of neighboring amateurs, the c -w transmitter MUST meet two important specifications. 1- It must have no parasitic oscillations either in the stage being keyed or in any succeeding stage. 2- It must have some device in the keying circuit capable of shaping the leading and trailing edge of the waveform. Both these specifications must be met be/ore the transmitter is capable of c-w operation. Merely turning a transmitter on and off by the haphazard insertion of a telegraph key in some power lead is an invitation to trouble. The two general methods of keying a transmitter are those which control the excitation to the keyed amplifier, and those which control the plate or screen voltage applied to the keyed amplifier. Key -Click Elimination Key -click elimination is accomplished by preventing a too -rapid make- and -break of power to the antenna circuit, rounding off the keying characters so as to limit the sidebands to a value which does not cause interference to adjacent channels. Too much lag will prevent fast keying, but fortunately key clicks can be practically eliminated without limiting the speed of manual (hand) keying. Some circuits which eliminate key clicks introduce too much time lag and thereby add tails to the dots. These tails may cause the signals to be difficult to copy at high speeds. Considerable thought should be given as to which stage in a transmitter is the proper one to key. If the transmitter is keyed in a stage close to the oscillator, the change in r-f loading of the oscillator will cause the oscillator to shift frequency with keying. This will cause the signal to have a distinct chirp. The chirp will be multiplied as many times as the frequency of the oscillator is multiplied. A chirpy oscillator that would be passable on 80 meters would be unusable on 28 Mc. c.w. Keying the oscillator itself is an excellent way to run into keying difficulties. If no key click filter is used in the keying circuit, the transmitter will have bad key clicks. If a key click filter is used, the slow rise and decay of oscillator voltage induced by the filter action will cause a keying chirp. This action is Location of Keyed Stage O IS Figure CENTER -TAP KEYING WITH CLICK 9 FILTER The constants shown above are suggested as starting values; considerable variation in these values can be expected for optimum keying of amplifiers of different operating conditions. It is suggested that a keying relay be substituted for the key in the circuit above wherever practicable. true of all oscillators, whether electron coupled or crystal controlled. The more amplifier or doubler stages that follow the keyed stage, the more difficult it is to hold control of the shape of the keyed waveform. A heavily excited doubler stage or class C stage acts as a peak clipper, tending to square up a rounded keying impulse, and the cumulative effect of several such stages cascaded is sufficient to square up the keyed waveform to the point where bad clicks are reimposed on a clean signal. A good rule of thumb is to never key back farther than one stage removed from the final amplifier stage, and never key closer than one stage removed from the frequency controlling oscillator of the transmitter. Thus there will always be one isolating stage between the keyed stage and the oscillator, and one isolating stage between the keyed stage and the antenna. At this point the waveform of the keyed signal may be most easily controlled. first place it may be established that the majority of new design transmitters, and many of those of older design as well, use a medium power beam tetrode tube either as the output stage or as the exciter for the output stage of a high power transmitter. Thus the transmitter usually will end up with a tube Keyer Circuit Requirements In the such as type 2E26, 807, 6146, 813, 4 -65A, 4E27/257B, 4 -125A or similar, or one of these tubes will be used as the stage just ahead of the output stage. Second, it may be established that it is undesirable to key further down in the transmitter chain than the stage just ahead of the final www.americanradiohistory.com HANDBOOK Transmitter Keying For 100 per cent protection, just obey the following rule: never work on the transmitter or reach inside any protective cover except when the green pilots are glowing. To avoid confusion, no other green pilots should be used on the transmitter; if you want an indicator jewel to show when the filaments are lighted, use amber instead of green. Filter capacitors of good quality hold their charge for some time, and when the voltage is more than 1000 volts it is just about as dangerous to get across an undischarged 4 -pfd. filter capacitor as it is to get across a high -voltage supply that is turned on. Most power supplies incorporate bleeders to improve regulation, but as these are generally wire -wound resistors, and as wire -wound resistors occasionally open up without apparent cause, it is desirable to incorporate an auxiliary safety bleeder across each heavy -duty bleeder. Carbon resistors will not stand much dissipation and sometimes change in value slightly with age. However, the chance of their opening up when run well within their dissipation rating is very small. To make sure that all capacitors are bled, it is best to short each one with an insulated screwdriver. However, this is sometimes awkward and always inconvenient. One can be virtually sure by connecting auxiliary carbon bleeders across all wire -wound bleeders used on supplies of 1000 volts or more. For every 500 volts, connect in series a 500,000 -ohm 1 -watt carbon resistor. The drain will be negligible (1 ma.) and each resistor will have to dissipate only 0.5 watt. Under these conditions the resistors will last indefinitely with little chance of opening up. For a 1500-volt supply, connect three 500,000 -ohm resistors in series. If the voltage exceeds an integral number of 500 volt divisions, assume it is the next higher integral value; for instance, assume 1800 volts as 2000 volts and use four resistors. Do not attempt to use fewer resistors by using a higher value for the resistors; not over 500 volts should appear across any single 1 -watt resistor. In the event that the regular bleeder opens up, it will take several seconds for the auxiliary bleeder to drain the capacitors down to a safe voltage, because of the very high resistance. Therefore, i t is best to allow 10 or 15 seconds after turning off the plate supply before attempting to work on the transmitter. If a 0 -1 d-c milliammeter is at hand, it may be connected in series with the auxiliary bleeder to act as a high voltage voltmeter. Safety Bleeders "Hot" Adjustments Some amateurs contend that it is almost impossible 391 to make certain adjustments, such as coupling and neutralizing, unless the transmitter is running. The best.thing to do is to make all neutralizing and coupling devices adjustable from the front panel by means of flexible control shafts which are broken with insulated couplings to permit grounding of the panel bearing. If your particular transmitter layout is such that this is impracticable and you refuse to throw the main switch to make an adjustment -throw the main switch -take a reading -throw the main switch-make an adjustment -and so on, then protect yourself by making use of long adjusting rods made from 1/-inch dowel sticks which have been wiped with oil when perfectly free from moisture. If you are addicted to the use of pickup loop and flashlight bulb as a resonance and neutralizing indicator, then fasten it to the end of a long dowel stick and use it in that manner. Protective Interlocks increasing tendency toward construcWith the tion of transmitters in enclosed steel cabinets a transmitter becomes a particularly lethal device unless adequate safety provisions have been incorporated. Even with a combined safety signal and switch as shown in figure 8 it is still conceivable that some person unfamiliar with the transmitter could come in contact with high voltage. It is therefore recommended that the transmitter, wherever possible, be built into a complete metal housing or cabinet and that all doors or access covers be provided with protective interlocks (all interlocks must be connected in series) to remove the high voltage whenever these doors or covers are opened. The term "high voltage" should mean any voltage above approximately 150 volts, although it is still possible to obtain a serious burn from a 150 -volt circuit under certain circumstances. The 150 -volt limit usually will mean that grid -bias packs as well as high voltage packs should have their primary circuits opened when any interlock is opened. 20 -4 Transmitter Keying The carrier from a c -w telegraph transmitter must be broken into dots and dashes for the transmission of code characters. The carrier signal is of constant amplitude while the key is closed, and is entirely removed when the key is open. When code characters are being transmitted, the carrier may be considered as being modulated by the keying. If the change from the no- output condition to full -output, or vice versa, occurs coo rapidly, the rectangular pulses which form the keying characters contain high- frequency components which take up www.americanradiohistory.com 390 THE Transmitter Keying and Control sary chances. However, no one is infallible, and chances of an accident are greatly lessened if certain factors are taken into consideration in the design of a transmitter, in order to protect the operator in the event of a lapse of caution. If there are too many things one must "watch out for" or keep in mind there is a good chance that sooner or later there will be a mishap; and it only takes one. When designing or constructing a transmitter, the following safety considerations should be given attention. For the utmost in protection, everything of metal on the front panel of a transmitter capable of being touched by the operator should be at ground potential. This includes dial set screws, meter zero adjuster screws, meter cases if of metal, meter jacks, everything of metal protruding through the front panel or capable of being touched or nearly touched by the operator. This applies whether or not the panel itself is of metal. Do not rely upon the insulation of meter cases or tuning knobs for protection. The B negative or chassis of all plate power supplies should be connected together, and to an external ground such as a waterpipe. Grounds It is not necessary to resort to rack and panel construction in order to provide complete enclosure of all components and wiring of the transmitter. Even with metal- chassis construction it is possible to arrange things so as to incorporate a protective shielding housing which will not interfere with ventilation yet will prevent contact with all wires and components carrying high voltage d.c. or a.c., in addition to offering shielding action. If everything on the front panel is at ground potential (with respect to external ground) and all units are effectively housed with protective covers, then there is no danger except when the operator must reach into the interior part of the transmitter, as when changing coils, neutralizing, adjusting coupling, or shooting trouble. The latter procedure can be made safe by making it possible for the operator to be absolutely certain that all voltages have been turned off and that they cannot be turned on either by short circuit or accident. This can be done by incorporation of the following system of main primary switch and safety signal lights. Exposed Wires and Components The common method of using red pilot lights to show when a circuit is on is useless except from an ornamental standpoint. When the red pilot is not lit it usually means that the circuit is turned off, but it can Combined Safety Signal and Switch RADIO 3V TO GREEN PILOT LIGHTS ON FRONT PANEL AND ON EACH CHASSIS 6 .Q MAIN 113 V. SUPPLY 0 -. 11 FIL TRANS -(D PDT SWITCH IS V.A C TO ENTIRE TRANSMITTER Figure 8 COMBINED MAIN SWITCH AND SAFETY SIGNAL When shutting down the transmitter, throw the main switch to neutral. If work is to be done on the transmitter, throw the switch all the way to "pilot," thus turning on the green pilot lights on the panel and on each chassis, and insuring that no voltage can exist on the primary of any transformer, even by virtue of a short or accidental ground. mean that the circuit is on but the lamp is burned out or not making contact. To enable you to touch the tank coils in your it is cept sers transmitter with absolute assurance that impossible for you to obtain a shock exfrom possible undischarged filter conden(see following topic for elimination of this hazard), it is only necessary to incorporate a device similar to that of figure 8. It is placed near the point where the main 110 -volt leads enter the room (preferably near the door) and in such a position as to be inaccessible to small children. Notice that this switch breaks both leads; switches that open just one lead do not afford complete protection, as it is sometimes possible to complete a primary circuit through a short or accidental ground. Breaking just one side of the line may be all right for turning the transmitter on and off, but when you are going to place an arm inside the transmitter, both 110 -volt leads should be broken. When you are all through working your transmitter for the time being, simply throw the main switch to neutral. When you find it necessary to work on the transmitter or change coils, throw the switch so that the green pilots light up. These can be ordinary 6.3-volt pilot lamps behind green bezels or dipped in green lacquer. One should be placed on the front panel of the transmitter; others should be placed so as to be easily visible when changing coils or making adjustments requiring the operator to reach inside the transmitter. www.americanradiohistory.com Safety Precautions HANDBOOK 389 113 VOLT SUPPLY FOR ENTIRE TRANSMITTER AT OPERATING POSITION TRANSMIT STOP USES RECEIVE I SAFETY SWITCH (SEE FIG.12) L-t-~y- PROTECTIVE -CL ---121aCcH INTERLOCKS OVERLOAD' CONTACTS ORECEIVER POWER Lt TRANSFORMER C.T. THERMAL TIME-DELAY RELAY HIGH VOLT. FILS. ON ,000, STANDBY (I (I13 V.) 13 V. ANTENNA CHANGEOVER RELAY TUNE-UP SWITCH INDICATOR LIGHTS 1000, IIII17 SW., ALL FILAMENT TRANSFORMERS EXCITER M.V. TRANSFORMER Figure fllllll HIGH VOLTAGE TRANSFORMER 7 PUSH -BUTTON TRANSMITTER -CONTROL CIRCUIT Pushing the START button either at the transmitter or at the operating position will light all filaments and start the time -delay r e I a y in its cycle. When the c y c l e has been completed, a touch of the TRANSMIT button will put the transmitter on the air and disable the receiver. Pushing the RECEIVE button will disable the transmitter and restore the receiver. Pushing the STOP button will instantly drop the entire transmitter from the a -c line. If desired, a switch may be placed in series with the lead from the RECEIVE button to the protective interlocks; opening the switch will make it impossible for any person accidentally to put the transmitter on the air. Various other safety provisions, such as the protective- interlock arrangement described in the text have been incorporated. With the circuit arrangement shown for the overload -relay contacts, it is only necessary to use a simple normally - closed d -c relay with a variable shunt across the coil of the relay. When the current through the coil becomes great enough to open the normally-closed contacts the hold circuit on the plate-voltage relay will be broken and the plate voltage will be removed. If the overload is only momentary, such as a modulation peak or a tank flashover, merely pushing the TRANSMIT button will again put the transmitter on the air. This simple circuit provision elimi- nates the requirement for expensive overload relays of the mechanically -latching type, but gives excellent overload protection. button momentarily to light the transmitter filaments and start the time -delay relay in its cycle. When the standby light comes on it is only necessary to touch the TRANSMIT button to put the transmitter on the air and disable the receiver. Touching the RECEIVE button will turn off the transmitter and restore the receiver. After a period of operation it is only necessary to touch the STOP button at either the transmitter or the operating position to shut down the transmitter. This type of control arrangement is called an electrically- locking push -to- still transmit control system. Such systems are frequently used in industrial electronic control. 20 -3 Safety Precautions The best way for an operator to avoid serious accidents from the high voltage supplies of a transmitter is for him to use his head, act only with deliberation, and not take unneces- www.americanradiohistory.com RADIO THE Transmitter Keying and Control 388 VOLT SUPPLY FOR ENTIRE TRANSMITTER 115 FUSES SAFETY SWITCH (SEE FIGS ) \St °RECEIVER POWER TRANSFORMER C.T. HUSKY TOGGLE SWITCH ON TRANSMITTER PROTECTIVE INTERLOCKS THERMAL TIME -DELAY RELAY O O TRANSMITRECEIVE SWITCH O HIGH VOLT. FI LS. STANDS (115V.) O11S V. ANTENNA CHANGEOVER O RELAY TUNE -UP SWITCH INDICATOR LIGHTS ,Qoo, ,000, ?-1-AMENT TRANSFORMERS ALL r1 -.000, 3V. ,Qoo) EXCITER M.V. TRANSFORMER .000, HIGH VOLTAGE TRANSFORMER Figure 6 TRANSMITTER CONTROL CIRCUIT Closing S1 lights all filaments in the transmitter and starts the time -delay relay in its cycle. When the time -delay relay has operated, closing the transmit -receive switch at the operating position will apply plate power to the transmitter and disable the receiver. A tune-up switch hos been provided so that the exciter stages may be tuned without plate voltage on the final amplifier. mister on the air, has had the experience of having to throw several switches and pull or insert a few plugs when changing from receive to transmit. This is one extreme in the direction of how not to control a transmitter. At the other extreme we find systems where it is only necessary to speak into the microphone or touch the key to change both transmitter and receiver over to the transmit condition. Most amateur stations are intermediate between the two extremes in the control provisions and use some relatively simple system for transmitter control. In figure 5 is shown an arrangement which protects mercury -vapor rectifiers against premature application of plate voltage without resorting to a time -delay relay. No matter which switch is thrown first, the filaments will be turned on first and off last. However, double pole switches are required in place of the usual single -pole switches. When assured time delay of the proper interval and greater operating convenience are desired, a group of inexpensive a -c relays may be incorporated into the circuit to give a control circuit such as is shown in figure 6. This arrangement uses a 115 -volt thermal (or motoroperated) time -delay relay and a d -p -d -t 115 volt control relay. Note that the protective interlocks are connected in series with the coil of the relay which applies high voltage to the transmitter. A tune -up switch has been included so that the transmitter may be tuned up as far as the grid circuit of the final stage is concerned before application of high voltage to the final amplifier. Provisions for operating an antenna- changeover relay and for cutting the plate voltage to the receiver when the transmitter is operating have been included. A circuit similar to that of figure 6 but incorporating push- button control of the transmitter is shown in figure 7. The circuit features a set of START -STOP and TRANSMIT-RECEIVE buttons at the transmitter and a separate set at the operating position. The control push buttons operate independently so that either set may be used to control the transmitter. It is only necessary to push the START www.americanradiohistory.com HANDBOOK Transmitter Control 387 TO EXCITER POWER SUPPLIES TO N.V. POWER SUPPLY C 115V.A.0 LINE ',INTERLOCKS Jt IN TRANSMITTER LINE PLUGI TRANSMITTER GREEN FILAMENT PILOT POWER CONTROL TRANSFORMERS RELAY PLUG FOR ABLE TO VARIAC OR 115 V. TO EXCITER AND HIGH.VOLTAGE RELAYS. AND TO RECEIVER CONTROL AND ANTENNA ROWERSTAT RED PI LOT DUMMY PLUG FOR STRAIGHT OPERATION L FILAMENT TRANSFORMERS TO EXTERNAL VARIAC OR POWERSTAT Figure 4 CIRCUIT WITH VARIABLE -RATIO AUTO -TRANSFORMER When the dummy plug is inserted into the receptacle on the equipment, closing of the power control relay will apply full voltage to the primaries. With the cable from the Variac or Powerstat plugged into the socket the voltage output of the high -voltage power supply may be varied from zero to about IS per cent above normal. One convenient arrangement for using a Variac or Powerstat in conjunction with the high -voltage transformer of a transmitter is illustrated in figure 4. In this circuit a heavy three -wire cable is run from a plug on the transmitter to the Variac or Powerstat. The Variac or Powerstat then is installed so that it is accessible from the operating desk so that the input power to the transmitter may be controlled during operation. If desired, the cable to the Variac or Powerstat may be unplugged from the transmitter and a dummy plug inserted in its place. With the dummy plug in place the transmitter will operate at normal plate voltage. This arrangement allows the transmitter to be wired in such a manner that an external Variac or Powerstat may be used if desired, even though the unit is not available at the time that the transmitter is constructed. Notes on the Use of the Variac or Powerstat Plate voltage to the modula - tors may be controlled at the same time as the plate voltage to the final amplifier is varied if the modulator stage uses beam tetrode tubes; variation in the plate voltage on such tubes used as modulators causes only a moderate change in the standing plate current. Since the final amplifier plate voltage is being controlled simultaneously with the modulator CHANGEOVER RELAYS 52 J Figure 5 PROTECTIVE CONTROL CIRCUIT With this circuit arrangement either switch may be closed first to light the heaters of all tubes and the filament pilot light. Then when the second switch is closed the high voltage will be applied to the transmitter and pilot will light. With a 30- second delay between the closing of the first switch and the closing of the second, the rectifier tubes will be adequately protected. Similarly, the opening of either switch will remove plate voltage from the rectifiers while the heaters remain lighted. the red plate voltage, the conditions of impedance match will not be seriously upset. In several high power transmitters using this system, and using beam -tetrode modulator tubes, it is possible to vary the plate input from about 50 watts to one kilowatt without a change other than a slight increase in audio distortion at the adjustment which gives the lowest power output from the transmitter. With triode tubes as modulators it usually will be found necessary to vary the grid bias at the same time that the plate voltage is changed. This will allow the tubes to be operated at approximately the same relative point on their operating characteristic when the plate voltage is varied. When the modulator tubes are operated with zero bias at full plate voltage, it will usually be possible to reduce the modulator voltage along with the voltage on the modulated stage, with no apparent change in the voice quality. However, it will be necessary to reduce the audio gain at the same time that the plate voltage is reduced. 20 -2 Transmitter Control Methods Almost everyone, when getting a new trans- www.americanradiohistory.com 386 Transmitter Keying TO EXCITER POWER SUPPLIES T s o o V.A.0 LINE 115 HI -LO POWER RELAY POWER CONTROL RELAY TO FILAMENT TRANSFORMERS o s 'Tv o 230 v. A C SINGLE PHASE WITH GROUNDED HI -LO K2 POWER K1 RELAY NEUTRAL POWER CONTROL RELAY TRANSFORMERS Figure 3 FULL -VOLTAGE /HALF -VOLTAGE POWER CONTROL SYSTEMS The circuit at (A) is for use with o 115 -volt a -c line. Transformer T is of the standard type having two 11S -volt primaries; these primaries are connected in series for half voltage output when the power control relay Kt is energized but the hi -lo relay K2 is not operated. When both relays are energized the full output voltage is obtained. At (B) is a circuit for use with a standard 230 -volt residence line with grounded neutral. The two relays control the output of the power sup- plies the THE and Control some as at (A). primaries in parallel will deliver full output from the plate supply. Then when the two primaries are connected in series and still operated from the 115 -volt line the output voltage from the supply will be reduced approximately to one half. In the case of the normal class C amplifier, a reduction in plate voltage to one half will reduce the power input to the stage to one quarter. If the transmitter is to be operated from a 230 -volt line, the usual procedure is to operate the filaments from one side of the line, the RADIO low- voltage power supplies from the other side, and the primaries of the high -voltage transformer across the whole line for full power output. Then when reduced power output is required, the primary of the high -voltage plate transformer is operated from one side to center tap rather than across the whole line. This procedure places 115 volts across the 230 -volt winding the same as in the case discussed in the previous paragraph. Figure 3 illustrates the two standard methods of power reduction with a plate transformer having a double primary; (A) shows the connections for use with a 115 -volt line and (B) shows the arrangement for a 230 -volt a -c power line to the transmitter. The full- voltage /half- voltage methods for controlling the power input to the transmitter, as just discussed, are subject to the limitation that only two levels of power input (full power and quarter power) are obtainable. In many cases this will be found to be a limitation to flexibility. When tuning the transmitter, the antenna coupling network, or the antenna system itself it is desirable to be able to reduce the power input to the final stage to a relatively low value. And it is further convenient to be able to vary the power input continuously from this relatively low input up to the full power capabilities of the transmitter. The use of a variable -ratio auto -transformer in the circuit from the line to the primary of the plate transformer will allow a continuous variation in power input from zero to the full capability of the transmitter. Variable -Ratio There are several types Auto- Transformers of variable -ratio auto- transformers available on the market. Of these, the most common are the Variac manufactured by the General Radio Company, and the Pouerstat manufactured by the Superior Electric Company. Both these types of variable -ratio transformers are excellently constructed and are available in a wide range of power capabilities. Each is capable of controlling the line voltage from zero to about 15 per cent above the nominal line voltage. Each manufacturer makes a single -phase unit capable of handling an output power of about 175 watts, one capable of about 750 to 800 watts, and a unit capable of about 1500 to 1800 watts. The maximum power- output capability of these units is available only at approximately the nominal line voltage, and must be reduced to a maximum current limitation when the output voltage is somewhat above or below the input line voltage. This, however, is not an important limitation for this type of application since the output voltage seldom will be raised above the line voltage, and when the output voltage is reduced below the line voltage the input to the transmitter is reduced accordingly. www.americanradiohistory.com Transmitter Control HANDBOOK not drop more than 5 volts (assuming a 117 volt line) under load and the wiring does not overheat, the wiring is adequate to supply the transmitter. About 600 watts total drain is the maximum that should be drawn from a 117 -volt lighting outlet or circuit. For greater power, a separate pair of heavy conductors should be run right from the meter box. For a 1 -kw. phone transmitter the total drain is so great that a 230 -volt "split" system ordinarily will be required. Most of the newer homes are wired with this system, as are homes utilizing electricity for cooking and heating. With a three -wire system, be sure there is no fuse in the neutral wire at the fuse box. A neutral fuse is not required if both "hot" legs are fused, and, should a neutral fuse blow, there is a chance that damage to the radio transmitter will result. If you have a high power transmitter and do a lot of operating, it is a good idea to check on your local power rates if you are on a straight lighting rate. In some cities a lower rate can be obtained (but with a higher "minimum") if electrical equipment such as an electric heater drawing a specified amount of current is permanently wired in. It is not required that you use this equipment, merely that it be permanently wired into the electrical system. Naturally, however, there would be no saving unless you expect to occupy the same dwelling for a considerable length of time. Outlet Strips The outlet strips which have been suggested for installation in the baseboard or for use on the rear of a desk are obtainable from the large electrical supply houses. If such a house is not in the vicinity it is probable that a local electrical contractor can order a suitable type of strip from one of the supply house catalogs. These strips are quite convenient in that they are available in varying lengths with provision for inserting a -c line plugs throughout their length. The a -c plugs from the various items of equipment on the operating desk then may be inserted in the outlet strip throughout its length. In many cases it will be desirable to reduce the equipment cord lengths so that they will plug neatly into the outlet strip without an excess to dangle behind the desk. Contactors and The use of power -control con tactors and relays often will add considerably to the operating convenience of the station installation. The most practicable arrangement usually is to have a main a -c line switch on the front of the transmitter to apply power to the filament transformers and to the power control circuits. It also will be found quite convenient to have a single a -c line switch on the operating desk Relays 385 to energize or cut the power from the outlet strip on the rear of the operating desk. Through the use of such a switch it is not necessary to remember to switch off a large number of separate switches on each of the items of equipment on the operating desk. The alternative arrangement, and that which is approved by the Underwriters, is to remove the plugs from the wall both for the transmitter and for the operating -desk outlet strip when a period of oper- ation has been completed. While the insertion of plugs or operation of switches usually will be found best for applying the a -c line power to the equipment, the changing over between transmit and receive can best be accomplished through the use of relays. Such a system usually involves three relays, or three groups of relays. The relays and their functions are: (1) power control relay for the transmitter -applies 115 -volt line to the primary of the high- voltage transformer and turns on the exciter; (2) control relay for the receiver -makes the receiver inoperative by any one of a number of methods when closed, also may apply power to the v.f.o. and to a keying or a phone monitor; and (3) the antenna changeover relay- connects the antenna to the transmitter when the transmitter is energized and to the receiver when the transmitter is not operating. Several circuits illustrating the application of relays to such control arrangements are discussed in the paragraphs to follow in this chapter. Controlling Transmitter Power Output It is necessary, in order to comply with FCC regulations, that transmitter power output be limited to the minimum amount necessary to sustain communication. This requirement may be met in several ways. Many amateurs have two transmitters; one is capable of relatively high power output for use when calling, or when interference is severe, and the other is capable of considerably less power output. In many cases the lower powered transmitter acts as the exciter for the higher powered stage when full power output is required. But the majority of the amateurs using a high powered equipment have some provision for reducing the plate voltage on the high -level stages when reduced power output is desired. One of the most common arrangements for obtaining two levels of power output involves the use of a plate transformer having a double primary for the high -voltage power supply. The majority of the high -power plate transformers of standard manufacture have just such a dual primary arrangement. The two primaries are designed for use with either a 115 -volt or 230 volt line. When such a transformer is to be operated from a 115 -volt line, operation of both www.americanradiohistory.com 384 __ - FROM LINE THE Transmitter Keying and Control TO OTHER HOUSE CIRCUITS RADIO SHORT CORDS FROM RECEIVER VF.O..CLOCR FRED METER, OUTLET STRIP. PLAN OA TO PLAN pB Figure 1 THE PLAN (A) POWER SYSTEM A -c line power from the main fuse box in the house Is run separotely to the receiving equipment and to the transmitting equipment. Separate switches and fuse blocks then are available for the transmitters and for the auxiliary equipment. Since the fuses in the boxes at the operating room will be in series with those at the main fuse box, those in the operating room should have a lower rating than those at the main fuse box. Then It will always be possible to replace blown fuses without leaving the operating room. The fuse boxes can conveniently be located alongside one another on the walla the operating room. type. It is possible also that the BX cable will have to be permanently affixed to the transmitter with the connector at the fuse -box end. These details may be worked out in advance with the electrical inspector for your area. The general aspects of Plan ( B) are shown in figure 2. The basic difference between the two plans is that (A) represents a permanent installation even though a degree of mobility is allowed through the use of BX for power leads, while plan (B) is definitely a temporary type of installation as far as the electrical inspector is concerned. While it will be permissible in most areas to leave the transmitter cord plugged into the outlet even though it is turned off, the Fire Insurance Underwriters codes will make it necessary that the cord which runs to the group of outlets at the back of the operating desk be removed whenever the equipment is not actually in use. Whether the general aspects of plans (A) or (B) are used it will be necessary to run a number of control wires, keying and audio leads, and an excitation cable from the operating desk Figure 2 THE PLAN (B) POWER SYSTEM This system is less convenient than the (A) system, but does not require extensive rewiring of the electrical system within the house to accommodate the arrangement. Thus it is better for a temporary or semi-permanent installation. In most cases it will be necessary to run an extra conduit from the main fuse box to the outlet from which the transmitter is powered, since the standard arrangement in most houses is to run all the outlets In one room (and sometimes all in the house) from a single pair of fuses and leads. to the transmitter. Control and keying wires can best be grouped into a multiple -wire rubber covered cable between the desk and the transmitter. Such an arrangement gives a good appearance, and is particularly practical if cable connectors are used at each end. High -level audio at a moderate impedance level (600 ohms or below) may be run in the same control cable as the other leads. However, low -level audio can best be run in a small coaxial cable. Small coaxial cable such as RG -58 /U or RG -59/U also is quite satisfactory and quite convenient for the signal from the v.f.o. to the r -f stages in the transmitter. Coaxial -cable connectors of the UG series are quite satisfactory for the terminations both for the v -f -o lead and for any low -level audio cables. Checking an To make sure that an outlet will Outlet with a stand the full load of the entire transmitter, plug in an electric heater rated at about 50 per cent greater wattage than the power you expect to draw from the line. If the line voltage does Heavy Load www.americanradiohistory.com CHAPTER TWENTY Transmitter Keying and Control 20 -1 Since the usual home outlet is designed to handle only about 600 watts maximum, the transmitter, unless it is of relatively low power, should be powered from another source. This procedure is desirable in any event so that the voltage supplied to the receiver, frequency control, and frequency monitor will be substantially constant with the transmitter on or off the air. So we come to two general alternative plans with their variations. Plan (A) is the more desirable and also the most expensive since it involves the installation of two separate lines from the meter box to the operating position either when the house is constructed or as an alteration. One line, with its switch, is for the transmitters and the other line and switch is for receivers and auxiliary equipment. Plan ( B) is the more practicable for the average amateur, but its use requires that all cords be removed from the outlets whenever the station is not in use in order to comply with the electrical codes. Figure 1 shows a suggested arrangement for carrying out Plan (A). In most cases an installation such as this will require approval of the plans by the city or county electrical inspector. Then the installation itself will also require inspection after it has been completed. It will be necessary to use approved outlet boxes at the rear of the transmitter where the cable is connected, and also at the operating bench where the other BX cable connccts to the outlet strip. Also, the connectors at the rear of the transmitter will have to be of an approved Power Systems It is probable that the average amateur station that has been in operation for a number of years will have at least two transmitters available for operation on different frequency bands, at least two receivers or one receiver and a converter, at least one item of monitoring or frequency measuring equipment and probably two, a v.f.o., a speech amplifier, a desk light, and a clock. In addition to the above 8 or 10 items, there must be an outlet available for a soldering iron and there should be one or two additional outlets available for plugging in one or two pieces of equipment which are being worked upon. It thus becomes obvious that 10 or 12 outlets connected to the 115-volt a -c line should be available at the operating desk. It may be practicable to have this number of outlets installed as an outlet strip along the baseboard at the time a new home is being planned and constructed. Or it might be well to install the outlet strip on the operating desk so as to have the flexibility of moving the operating desk from one position to another. Alternatively, the outlet strip might be wall mounted just below the desk top. of all the items of equipment, other than transmitters, used at the operating position is totalled, you probably will find that 350 to 600 watts will be required. Power Drain Per Outlet When the power drain 383 www.americanradiohistory.com HANDBOOK Deluxe Mobile Transceiver A complete chassis -assembly mock -up should be made up of cardboard sheets, and the various parts laid out in order to ascertain their final position. The tuning capacitor gang is made up of two dual units and two single units, with their shafts cut to length so that the over-all depth of the gang allows room for the p.a. plate coil and associated padding capacitors. done this type of assembly and wiring as a vocation will find this style of construction interesting and a challenge to his ingenuity. The beauty of the final equipment is well worth the time and study it takes to design and lay out a unit of this order of complexity. Transceiver Alignment and Test Transceiver The under -chassis wiring may Wiring be observed in figure 38. All power wiring is laced to form a harness that runs about the chassis in a square loop centered about the coil assembly. Small components are mounted directly to tube socket pins, to lug terminal strips, or to small phenolic terminal boards. Ground connections are made to lugs placed beneath socket retaining bolts. The r.f. components of the receiver occupy the center portion of the chassis. Small inter stage shields made of durai separate the r.f., mixer, and oscillator stages, and an additional shield plate covers the bottom of the 6AH6 v.f.o. compartment. To the rear of this compartment are the driver stages of the transmitter section. A wiring harness of the type used in this transceiver may best be made up external to the unit. A layout of the harness and the terminations of the various wires is sketched full -size on a large board and the wires are then laid out on the board in their proper positions, cut to length, and laced. The completed harness is then dropped into the equipment and the terminations made. An amateur experienced in equipment construction, or who has TRANSCEIVER Amplifier tuning and loading controls are mounted on rear of the cabinet. Below (left to right) are: antenna receptacle, power receptacle, speaker receptacle, and 5 -meter zero -set po- tentiometer. Additional ventilation is provided by rows of holes across rear of cabinet. 11 _ t!! tft t t! When the transceiver is completed, all wiring should be checked and "rung out" to preclude the possibility of wiring errors or accidental grounds. The tubes are now placed in the unit, and the various tuned circuits adjusted to their approximate operating range by means of a grid-dip oscillator. The transmitter and modulator tubes are removed, and the receiver section is aligned in the following manner: The first step is to align the low frequency i.f. strip. A low level modulated 260 kc. signal is injected into the plate circuit of the 6BE6 second mixer and transformers T.., T3, and T4 are adjusted for maximum receiver output. Next, oscillator coil L of the 6BE6 stage is adjusted for maximum #1 grid current and a 4.26 Mc. signal is fed to the input circuit of the 6BA7 first mixer. Transformer Tt is adjusted for maximum signal strength. A 29 Mc. signal is now applied to the antenna circuit of the receiver, and the main tuning dial is adjusted to this approximate setting. Coil L3 and capacitor Ca of the master oscillator are adjusted until the test signal is heard. The tuned circuit of the oscillator is aligned to cover the span of 23,740- 25,440 kc., with equal leeway on each end of the range. The test signal is now placed on 29.5 Mc. and the padding capacitors of the r.f. and Figure 39 REAR VIEW OF 563 www.americanradiohistory.com tt 564 THE RADIO Receivers and Transceivers mixer stage are adjusted for maximum signal. The signal is next shifted to 28.5 Mc. and the variable slugs of the r.f. and mixer coils are adjusted in turn. This process is repeated until the tuning range of the receiver is correct, and the r.f. stages track properly across the dial. The transmitter section may now be aligned. The tubes are inserted in their sockets and relay RY1 is activated. The screen power lead to the 6146 is temporarily opened to disable that stage. Once again, the transceiver is tuned to 29.5 Mc. and the two padding capacitors of the 6CL6 buffer stages are adjusted for maximum grid drive to the 6146 stage. (Note: Grid current to the 6146 should be held to less than 4 ma. at all times) . The dial is now returned to 28.5 Mc. and the variable slugs of the buffer circuits are adjusted for maximum grid drive. The adjustments are repeated until reasonably constant grid drive occurs across the tuning range. The buffer stage and power amplifier are neutralized and screen voltage is applied to the 6146 tube. The transmitter frequency is set to 29.0 Mc. and the amplifier is tuned and loaded by means of the controls on the rear of the cabinet (figure 39) . The frequency of the transceiver is shifted to 29.5 Mc. and (without adjusting the loading capacitor) resonance is again reestablished with the rear tuning capacitor, Now, the frequency is shifted to 28.5 Mc. and auxiliary capacitor C14 is adjusted for resonance. This sequence of adjustment is repeated until proper resonance and loading occurs across the dial. Resonant plate current should be approximately 110 milliamperes and grid current should be 2 to 3 milliamperes. Modulator resting plate current is 25 milliamperes, rising to about 80 milliamperes under full modulation. C. The transmitter may be bench -tested with an a.c. power supply and a dummy load before it is placed in the automobile. Car mounting is accomplished by means of two heavy aluminum rails bolted to the top of the transceiver case which slide into suitable clamps affixed under the dash of the automobile as shown in figure 31. A transistor -type power supply or a dynamotor may be used. 250 volts at about 150 milliamperes, and 500 -600 volts at 200 milliamperes are required for operation of the transceiver. 27 -7 A Deluxe Receiver for the DX Operator The need exists for a high performance receiver, suitable for s.s.b., a.m., and c.w. operation that can be built in the home workshop at a modest price. The receiver should have a high order of stability and sensitivity and must have sufficient dynamic range to protect it against excessive cross- modulation caused by strong nearby signals. In addition, it should be possible to build the receiver without the use of special metal- handling tools. The receiver described in this section was designed to fill this need. It is a double conversion superheterodyne, employing crystal control in the first conversion stage and a tunable low frequency i.f. and mixer. This configuration provides maximum stability and permits the use of a dial calibrated directly in frequency. Collins mechanical filters and a Q- multiplier are used in the 455 kc. second intermediate frequency amplifier to provide the ultimate in selectivity and rejection and a product detector is employed for c.w. and Figure 40 FRONT VIEW OF DELUXE AMATEUR COMMUNICATION RECEIVER band receiver covers 80 -10 meters, with extra bond for 15 Mc. reception of WWV standard frequency signals. Collins mechanical filters provide ultimate in selectivity for s.s.b. a.m. phone, and c.w. The receiver employs a crystal controlled first conversion oscillator for high -order stability and "hang- a.g.c." for improved sideband reception. A simplified product detector is used for s.s.b. and c.w. operation. The precision dial can be read to one or two kilocycles on all bands. Room is provided above main dial for inclusion of v.h.f. converters for 2 and 6 meter operation, if desired. Six www.americanradiohistory.com HANDBOOK V2 V1 V4 IST MIX. R DX Communication Receiver 9 MC. Ve VS 2ND MIX. -OSC. 1ST I.F. (2.4 -2 2ND I.F. (33 KC.) ) 565 V7 3RD IF. 1--15.0 XC. FILTER VIAL OSC. VS RECEIVER TUNING (oSC1LLAFOA 1.945-2.445 MC. X (SEE PIG. 42) ) VII Ve AM-CW-55B 2ND AUDIO VIS V14A VOLT. REGULATOR V,4e OSC. 5 -METER 100 RC 12AU7 ACC 12AU7t, Figure 41 BLOCK DIAGRAM OF DELUXE AMATEUR COMMUNICATION RECEIVER reception. An automatic gain control circuit (a.g.c.) is provided for sideband, and auxiliary equipment includes an S -meter and 100 kc. crystal calibrator. Reception of the 15 Mc. Standard Frequency (WWV) signal is incorporated for receiver calibration purposes. Construction is simplified by making the receiver in modules that may be built and tested one at a time. s.s.b. The Receiver diagram of the receiver circuit is shown in figure 41. Fourteen tubes are used, plus a voltage regulator. The power supply utilizes semiconductors to reduce heating effects. The R.F. Section. The receiver covers the amateur bands between 10 and 80 meters, with an extra bandswitch position for "spot" reception of WWV at 15 Mc. The r.f. stage employs a 6DC6 semi -remote cutoff pentode to provide maximum freedom from crosstalk and front -end overload. A triode -connected 6AH6 serves as a low noise mixer stage, with local oscillator injection on the control grid. The first conversion oscillator is crystal controlled using a 6BJ6 in a "hot cathode" circuit operating on the low frequency side of the received signal. Circuit A block Receiver tuning is accomplished at the first intermediate frequency range of 2.4 -2.9 megacycles. Each tuning range thus covers 500 kilocycles. Any 500 kc. segment of the 10 meter band may be utilized by the proper choice of the conversion crystal. The tunable portion of the receiver consists of a 6BJ6 i.f. amplifier, and a 6BE6 second mixer stage. The oscillator portion of the 6BE6 tube tunes the region 1.945 -2.455 Mcs. to provide a 455 kc. intermediate frequency. Both oscillators are voltage regulated for maximum frequency stability. The I.F. Section. Two i.f. stages are employed to provide sufficient receiver gain. The first stage uses a 6AH6 which directly follows the mechanical filters and the Q- multiplier circuit. The filters allow a choice of 0.5 kc. passband for c.w., or a 3.0 kc. passband for sideband. A.m. reception may be done by listening to one of the two sidebands, or a 6.0 kc. bandwidth filter may be substituted for the 3.0 kc. unit. The Q- multiplier places a rejection "notch" at any point in the filter passband to eliminate heterodyne interference. The depth of notch can be adjusted by an auxiliary control. The over -all gain of the receiver is set by adjusting the "r.f. gain" control which fixes the operating bias on the low frequency i.f. www.americanradiohistory.com 2 v N VI W ^ t O. 0/1 o d C O d o l Q O - u i V et Z C Y 0 é _0,.o '3 M N O m u l im`I I 6.l Mkv r (=-{n I o ---i1! - sY-- - n 00rO1 I ~ I o o ó tV C o Q W d 0, ii =t NWI LL N G p V p o M ce Y O ><' 1-2. J V Hn p Ó Q K C «of Y G « = Y VI f :E = p uN y C Y .Y íf- w X jis iI < m iV k 3 4 4 O O O N i 3 O www.americanradiohistory.com 1" iS 1000 n - O CQ E ` *"1 2C ó é Ó d tili - á _V o_ Q c È«o. 0 00600 r- 0 ÿ k `o a « G O 3 á DX Communication Receiver Figure 42 page) (See opposite -50 Aµfd. National UM -50 or equivalent UM -100 or equivalent C} F. Johnson SMB11, with 240 µµtd. silver mica shunted across each section Ci -22 gµl& silver mica capacitor with 7 -3S µµI& ceramic trimmer connected in parallel C, -120 µµtd. silver mica capacitor with 7-35 µµfd. ceramic trimmer connected in parallel RFC mh. J. W. Miller Co. 1:J300-1000 S,4,n,e,n- Centralab PA -305 assembly with 6 -inch shalt and six Centralab PA -17 ceramic sections (60 degree index) assembly with 4 -inch SSA,n,e --- Centralab PA -301 shaft and two Centralab PA -0 ceramic sections S3- Corner Plates of C, bent to short out filter T1, T2 W. Miller Co. #B -727RF coil with S -27 shield Lr W. Miller Co. re-727C coil with S -27 shield X, Crystals -International Crystal Mfg. Co. Dial- Eddystone. Available from British Radio Electronics, Ltd., 1833 Jefferson Place, N.W., Washington 6, D. C. All bypass capacitors are .01 ;lid., disc ceramic, 600 volt. High frequency oscillator capacitors are silver mica Mechanical filters- Collins Radio Co., 455 kc., style K C1-C4 C5-100 µofd. National -E. -1 -J. -J. stages and also on the tunable i.f, stage. The front end of the receiver operates at maximum sensitivity and gain at all times in order to override the inherent tube noise level of the various mixer stages. The Detector and Audio Section. A 6BE6 mixer tube is employed as a hybrid detector. For sideband and c.w. operation, it functions as a product detector, with injection on the #1 grid from the beat oscillator and signal injection on the #3 grid. For a.m. service, the beat oscillator is disabled, and the signal is switched to the #1 grid. Thus one tube serves two functions, and does both of them well. The beat oscillator is a 6BJ6, with variable injection taken from the plate circuit. The oscillator frequency may be moved across the passband of the i.f. system to provide a choice of upper or lower sideband reception, as desired. The automatic gain control system employs a separate 6BJ6 i.f. amplifier stage driving a simple "hang- a.g.c." system of the type described by W1DX in the January, 1957 issue of QST magazine. The 6BJ6 stage isolates the b.f.o. from the a.g.c. system and prevents oscillator voltage from leaking into the a.g.c. circuit. The latter circuit is especially designed for s.s.b. and c.w. reception. It has a very rapid response that prevents receiver overload 567 on a syllabic burst of s.s.b., instantly reducing receiver gain to prevent overloading. The gain reduction remains in effect as long as the signal is in evidence, then "hangs" on for about 0.5 second after the removal of the signal. This sequence of action reduces to a minimum the usual "thump" that occurs at the start of a syllable and removes the "rush" of background noise at the end of a syllable that occurs with a conventional a.v.c. system. A triple diode 6BC7 and one -half a 12AU7 double triode comprise the complete "hanga.g.c." system. The double diode system following transformer T., and the 470K/0.01 µfd. R -C network determine the "on" time of the "attack" system, permitting the 0.1 µfd. a.g.c. capacitor to charge up in a relatively quick time. The capacitor remains charged, as the 12AU7 triode is cut off by this action, and there remains no discharge path to ground in the a.g.c. circuit, even when the voltage across the "attack" R -C network is removed. The time constant of the "release" network is considerably longer, and after a predetermined period, the a.g.c. voltage across this network decays sufficiently to permit the triode section to conduct and discharge the a.g.c. line capacitor. The proper ratio of voltages in the two R -C circuits can easily be established by proper adjustment of transformer T. A slight degree of delayed a.g.c. action is provided by applying fixed bias to the "attack" diode to prevent the circuit from being tripped by back,.grc und noise or weak signals. The S- Meter, Audio System and Power Supply. The S -meter circuit is a simple vacuum tube voltmeter that compares the a.g.c. voltage against a fixed reference voltage. The circuit is balanced for a meter null with no signal input to the receiver, and a.g.c. voltage unbalances the circuit causing a reading on the meter placed in the bridge of the circuit. The meter may be used for all modes of reception, providing usable readings on c.w. signals as well as sideband or a.m. A single 6AK6 provides sufficient audio for earphone reception, or to drive a speaker to good room volume. Ignition and other pulse -type noise is effectively reduced by means of a peak noise clipper made up of two inexpensive semiconductor diodes. The power supply is a voltage doubler type utilizing inexpensive silicon rectifiers. High www.americanradiohistory.com ¡00000 1 -14 _Z.i ' .. -_ ; www.americanradiohistory.com DX Communication Receiver 569 voltage is regulated by an 0A2 for the entire receiver, and standby is accomplished by breaking the B -plus line from the supply. Three separate filament windings on the power transformer provide sufficient capacity to power all the tubes. The use of 150 milliampere filament tubes wherever possible reduces the filament drain considerably. The whole receiver runs reasonably cool because of the low plate voltage and choice of low filament power tubes, achieving a high order of thermal stability in a short period of time. Receiver Construction Figure 44 TUNABLE I.F. SECTION OF RECEIVER The tunable i.f. section cf the receiver is built upon a 3" x 5" x 7" aluminum chassis. Input and output connections are made via "phono- type" coaxial fittings and lengths of RG -58 U coaxial line. Tube in foreground is 68E6 mixer (Vs), and tube in the rear is 6816 tunable i.f. (V4). Ceramic padding capacitors C, (two) and Cv are mounted at right of chassis, with the three Li. coils atop the A receiver such as this is a complex device and its construction should only be under- taken by a person familiar with receiving equipment and who has built equipment of this category before. The receiver is built upon an aluminum chassis measuring 1531" x 11" x 3" in size, and is contained within a ventilated cabinet measuring 16" x 111'4" x 91 ') ". The tunable i.f. system is built as a separate unit on an aluminum chassis -box measuring 3" x 5" x 7" (figures 44 and 45). The mechanical filter assembly is also built as a separate unit in an aluminum box measuring 2" x 3" x chassis. Figure 45 UNDER -CHASSIS VIEW OF TUNABLE I.F. SECTION OF RECEIVER Tunable i.f. stage is isolated from second mixer by shield partition across middle of chassis box. Mixer and oscillator sections of Vs are separated by a small partition. Tuning capacitors are mounted to the shield partitions and are driven through metal shaft couplings. Power receptacle is at rear of chassis. Complete assembly is fastened to main chassis by six sheet metal screws. www.americanradiohistory.com 570 THE RADIO Receivers and Transceivers 51/4" ( figure 47) . The b.f.o. assembly is built within an aluminum box measuring 11/2"" x 2" x 23/ " (figure 48). The remainder of the receiver is built upon the main chassis. No receiver is better than its tuning dial, so the very excellent Eddystone geared slide rule dial is used. The dial is centered hori- zontally on the panel and vertical placement is adjusted so that the drive engages the shaft of the variable tuning capacitors of the tunable i.f. system. Figure 46 UNDER -CHASSIS VIEW OF COMMUNICATIONS RECEIVER The receiver is built in sections which may be checked out one at a time for sake of sim- plicity. Crystal controlled r.f. section is at left, with coil slugs projecting from front and back of assembly. Conversion crystals are mounted in holders on front partition. Near center of chassis is box containing mechanical filters and switch (figure 47). At right is partition holding Q- multiplier coil and potentiometer, with auxiliary notch control located on the panel. The product detector switch is driven off- center by two flexible couplings. Power supply, diode rectifiers and filter section are at lower right, with audio stages across bottom of chassis. The chassis, panel, and tunable i.f. chassis should be assembled and studied before any chassis holes are drilled. The dial cut -out should next be made, making sure of alignment of the dial with the variable capacitors. Placement of the remainder of the components is not at all critical. The Tunable I.P. System. It is best to construct this item first, as it determines dial position and placement of other major parts. A close -up of this assembly is shown in figures 44 and 45. The three variable capacitors are ganged by means of brass shaft bushings. The first capacitor is mounted to the front wall of the chassis -box, and the other two are placed on aluminum interior partitions. The 6BE6 mixer tube is mounted to the front with the 6BJ6 at the rear. Power connections are made to a miniature connector on the rear of the chassis, and input and output terminations are made through "phono- type" coaxial connectors and short lengths of RG -58/U coaxial line. The Mechanical Filter Assembly. A partition separates the input and output circuits of the filter assembly, as shown in figure 47. Bulk- www.americanradiohistory.com HANDBOOK DX Communication Receiver head mounting filters are used to achieve a maximum degree of isolation across the filter. The individual segments of the selectivity switch (S_) are mounted in each compartment, with the switch mechanism passing through the bulkhead. A spring wiping contact is made for the rotor arm of the switch, grounding it at the center bulkhead to prevent a leakage path around the filter from being formed. Input and output terminations are made via "phono- type" coaxial fittings and RG -58/U coaxial line. The input and output circuits of the filters must be tuned to frequency. This is accomplished by a 50 µµtd. variable padding capaFigure 47 INTERIOR VIEW OF MECHANICAL FILTER ASSEMBLY BOX Bulkhead mounting mechanical filters are mounted to interior partition which isolates input and output sections. Drive shaft of selectivity switch is grounded at point it passes through partition by a wiping spring to achieve maximum circuit isolation. Input and output tuning capacitors of filters are made up of SO ppld. variable ceramic trimmers connected in parallel with 75 µpfd. silver mica capacitors. Trimmers are adjusted for maximum signal response, in same manner as i.f. transformer capacitors. 11 571 citor placed across each circuit and adjustable from the bottom of the receiver. The R.T. Assembly. The r.f. assembly is constructed within the main chassis as shown in figure 50. The sockets for the 6DC6 r.f. stage, the 6AH6 mixer, and the 6BJ6 oscillator are mounted on the main chassis and the associated coils, tuning capacitors, and bandswitch are mounted to four vertical partitions fixed beneath the chassis. Slug -tuned coils are used for all circuits and are mounted in a horizontal position about the bandswitch. The r.f. and mixer coils can be aligned by means of a "TV- type" screwdriver thrust through holes in the rear of the chassis, while the r.f. coils are adjusted from the front of the assembly by means of a short screwdriver. The partitions are mounted so that a space of 2" exists between them, and the associated tube socket falls in the center of each space. The switch assembly passes through the partitions and, in fact, holds them in position by virtue of the switch arms and spacers. The individual switch segments are placed so that they are near the end of each coil. This results in a very compact assembly having extremely short leads to all coils. The coils are staggered about the circumference of a circle so that both the r.f. and mixer slugs can be reached from the rear www.americanradiohistory.com 572 THE RADIO Receivers and Transceivers without interference between the coils. The four partition plates are cut from 1/32 -inch aluminum stock, and follow the layout of figure 51. They are not notched at first. Rather, a cardboard template is cut out and marked for drilling as shown. Then all four partitions are clamped together and drilled along with the template. Corner notches are now cut and all edges filed so that all four partitions are as identical in size and shape as possible. Only the holes shown in figure 51 are common to all pieces. The front and rear partitions have other holes i.e., crystal sockets, antenna input, power lead holes, etc. These may be drilled during layout and assembly of the unit as required. The 1/2-inch flanges are then bent over, taking care to bend the front and rear pieces in the proper direction. - The coils should be wound to the data of figure 49, before the unit is assembled. Only three coils are used in the oscillator section as an r.f. choke is employed on the 80 and 40 meter bands. The 14 Mc. coils are jumpered across the switch and used for the WWV position on 15 Mcs. All coils should be wired to the bandswitch before the tuning Figure 48 REAR VIEW OF RECEIVER Placement of major parts may be seen in this view. B.f.o. components and tube are mounted in small aluminum box next to the front panel (left), with S -meter above main tuning dial. At right on panel is standby control switch, with noise limiter switch beneath it. Power transformer is at left rear of chassis. On rear apron of chassis are placed !1. to r.): 115 volt power receptacle, utility socket, break -in gain control, S -meter adjust potentiometer, speaker terminals, and coaxial antenna receptacle. At extreme right are pass through holes to permit alignment of high frequency r.f. coils. www.americanradiohistory.com HANDBOOK DX Communication Receiver 573 Coil Table Figure 49 Band L1 3/16" 1. closewound 6t #24e 40 3/16" 3/16" 1. 3/16" closewound closewound 33t #24e 12t #24e closewound L5 RFC RFC 16t #24e 8t #24e 40t #30e spaced length closewound closewound (11600 kc.) 14t #24e 7t #24e 23t #24e spaced as above closewound spaced to cover form (18,600 kc.) 3t #24e 10 20t #30e of form 3-1/2t #24e 3/16" 1. 15 55t #30e closewound 1. 4t #34e 20 L3 Ls, L4 9t #24e 80 1. 12t #24e 6t #24e 16t #24e spaced as above closewound spaced to cover form (25,600 kc.) All coils wound on XR -SO forms. Ls, L. wound first -then a loyer of 1/2" Scotch No. 33 tape wound on cold ends of L2, L4 coils and LI, L3 primaries wound over tape. Small strip of tope plus coil cement secures the free ends of LI, L3. 80 M coils L2, L4 have SOµ,.fd. padders soldered across terminals. capacitors are finally mounted in place. The last step is to use the unit as a template to mark the clearance holes on the rear of the chassis, which are drilled before the unit is finally installed in the chassis. Receiver Wiring. The remainder of the receiver wiring is simple and straightforward. The sideband -a.m. switch (S4) is offset from the panel hole to clear the Q- multiplier coil (Ls) mounted on an L- shaped bracket beneath the chassis (figure 46) The audio low pass filter coil (L9) is placed between the 6BE6 detector and 12AU7 audio socket. Long runs of a.c. leads are done in shielded wire, as are audio leads. . The receiver may be aligned in sections. The first step is to align the i.f. system and beat oscillator. Next, the tunable i.f. stages should be aligned and tracked. Finally, the r.f. sections are properly tuned. The i.f. system should be aligned to the center of the passband of the narrowest -bandwidth mechanical filter. In the case of the 500 cycle filter, the center frequency must be 455.0 kilocycles with very little tolerance. The Receiver Alignment system may be roughly aligned with the aid of an external signal generator coupled into the #3 grid of the 6BE6 second mixer. A 455 kc. signal of low amplitude is injected into the input circuit and the tuning capacitors across the filter terminals, plus transformers T3 and T4 are adjusted for maximum response. The Q-multiplier should be out of the circuit for this test (switch S3 closed). Care should be taken not to overload the i.f. system during alignment, so a relatively weak signal should be used for this portion of the adjustment. A.g.c. transformer T5 should then be adjusted to provide the proper "attack" and "release" time for the gain control circuit. Finally, the slug of the b.f.o. coil (L10) is set to place the beat oscillator signal at the center of the i.f. passband with the b.f.o. panel control set at mid- scale. The signal generator is now shifted to the input circuit of the 6BJ6 tunable i.f. stage. The main tuning dial is set at 500 (minimum circuit capacitance). The generator is adjusted to 2.90 Mc., and padding capacitor Cy of the oscillator section is adjusted for signal response. 1.f. and mixer padders C. are then tuned for maximum signal. At a dial reading www.americanradiohistory.com 574 THE RADIO Receivers and Transceivers Figure 50 UNDER -CHASSIS VIEW OF R.F. COIL ASSEMBLY The high frequency coils are placed in a circle about the bandswitch (figure 51). Coils and capacitors are mounted on four shield partitions which are located between the tube sockets. R.f. stage socket (V1) is at rear of chassis, with mixer socket (V2) in center, and crystal oscillator socket (V3) nearest the panel. Oscillator crystals are mounted on front partition. Entire assembly is shielded by aluminum cover plate. e of zero (maximum circuit capacitance), the tunable stages should resonate at 2.40 Mc. Attention should now be given to the front end stages. It is a good idea and a time saver to peak circuits L1 -L2 and L3 -L4 to the proper frequency with the aid of a grid -dip oscillator. Coil L; is adjusted for proper crystal oscillator operation, which may be monitored in a nearby receiver. The signal generator is now set to the center frequency of the 500 kilocycle band in use and a moderate signal is injected into the antenna circuit of the receiver. The main tuning dial is adjusted to receive the signal, and the r.f. and mixer coils are peaked for maximum response with the r.f. tuning capacitors set at mid -scale. Once alignment has been completed, the operator should familiarize himself with receiver operation. The last step is to adjust the "break -in" gain control so that the receiver may be used to monitor c.w. transmissions. The Figure 51 R.F. ASSEMBLY PLATES Four assembly plates are required, as shown. Each plate is drilled as necessary for mounting of small components, etc. PELATIvE POSiT,ON WHEN A5SEMBLED OF TUBES OR UM-50 -50 FORM OUTLINE OuTLINE 12 1 R TB B_ TO -- ITCH STATOR TAB L www.americanradiohistory.com -6 Di NOTE OWL - X ROTOR E HOLES 1 TAB e -p \ (MAO V ) i USED PER STAGE. SPACED EVERT 50. ON SHOWN NERE z -05WoTCNMOUNT SCREW BOA 5 121 2 HANDBOOK short across the control circuit is removed from the utility socket on the rear apron and the control adjusted for the desired standby sensitivity level. The control may be shorted out by an external switch or relay during periods of reception. Ti 6. 3 V., 4. 2 5 I.F. A ., T O (CREENLEADS) Figure 52 63V,4.0 SCHEMATIC, POWER SUPPLY FOR RECEIVER -4.5 H at 200 ma. Stancor C-1411 S, A -2 pole, 3 position rotary switch SRI, -200 ma. rectifier. Sarkes- Tarzian A.,TO I AUDIO U D O I R. F. (BROWN LEADS) 6.3 V., 2 OA TO TUNABLE (YELLOW LEADS) L. L11 SR, , 2 575 DX Communication Receiver M- 500 with dual mounting kit T7-117 volts at 200 ma. Three 6.3 volt windings at 2.0, 4.0, and 4.23 amperes, respectively. Stancor P -8158 r STa OFF sreY. oN-0 a+ 350 + 40 5W www.americanradiohistory.com I. 200 E & E TECHNI -SHEET CONVERSION TABLE MICRO = - (µ) ONE -MILLIONTH MILLI = (m) ONE -THOUSANDTH TO CHANGE FROM UNITS MICRO -UNITS MILLI -UNITS KILO -UNITS KILO (K) ONE THOUSAND (M) ONE MILLION MEGA TO OPERATOR MICRO -UNITS MILLI -UNITS X 1,000,000 X 1,000 KILO -UNITS MEGA -UNITS ± ± MILLI -UNITS UNITS MICRO-UNITS UNITS MEGA -UNITS UNITS MEGA -UNITS UNITS OF MEASUREMENT KILO -UNITS UNITS - ± X X 1,000 or X 1,000,000 or X or 108 or 103 10 -3 10 -8 1,000 or X 10 -3 1,000,000 or X 10 -6 X 1,000 or = 1,000 or - 1,000 or X 1,000 or X X 103 X X 10 -3 X 1,000 or X 1,000,000 X or X www.americanradiohistory.com 10 -3 103 103 108 CHAPTER TWENTY -EIGHT Low Power Transmitters and Exciters The transmitter is the "heart" of the amateur station. Various forms of amplifiers and power supplies may be used in conjunction with basic exciters to form transmitters which will fit almost any requirement. Several different types of transmitting equipment designed to meet a wide range of needs are outlined in this chapter. A simple transistorized transmitter for 50 Mc. is described. This unit is a good introductory project for the amateur to "cut his teeth on" relative to the field of transistors. Also shown is a complete, TVIproof, medium- powered all -band phone and c.w. transmitter. A "W9TO" electronic keyer is illustrated, together with newly -developed "Strip Line" circuits which are applicable to the v.h.f. spectrum. For the amateur who is interested in the construction phase of his hobby, these units should offer interesting ideas which might well fit in with the design of his basic transmitting equipment. Figure 1. A POCKET -SIZE 50 MC. TRANSISTORIZED PHONE TRANSMITTER. Capable of 100 milliwatts input, this "collector modulated" six meter phone transmitter will provide amazing results when used with a good antenna system. The complete unit may be held in the palm of the hand. Panel controls are 11. to r.): crystal oscillator tuning (top) and audio gain control (bottom), multi meter, amplifier tuning (top) and loading (bottom). Switch on left is the multi -meter switch, with power switch at right. Microphone plug is centered between switches. www.americanradiohistory.com 578 THE RADIO Low Power Transmitters RCA RCA 2N3B4 2N384 5O AK X1 (PNP) L2 (P)(P) LI SO R ADJUST !/AS t OSC w--AMP 2 N44 (PNP) MIC ALL RESISTORS 1/2 WATT. 15V. -ISV AUX. Figure 2. SCHEMATIC, 50 MC. TRANSISTORIZED TRANSMITTER. L1-6 turns =18 wire, 58 inch diameter, S/8 inch long. (B8W miniductor 3007.) Top three turns from transistor end L2, L3 -Make both coils from a single piece of 88W miniductor =3007. Use nine turns. Cut coil between sixth and seventh turn, making two coils having six and two turns, 28 -1 A Transistorized 50 Mc. Transmitter and Power Supply The simple 50 Mc. transistorized transmitter shown in this section makes an interesting project for the amateur who wishes to familiarize himself with high frequency transistors. Capable of 100 milliwatts input, this little phone transmitter will give a good account of itself when it is used in conjunction with a beam antenna. It may be run from batteries or from a regulated a.c. power supply. Circuit The transmitter circuit utilizing inexpensive PNP -type transistors is shown in figure 2. The oscillator is crystal controlled, employDescription respectively, separated by a distance one turn M -10 ma., d.c., 11/4" square meter -0 T1- Transistor of transformer, SK to 80K. Thor - darson TR -13 T2, Ta-Transistor transformer, 10K to 2K. Triad TY -56X ing a 2N384 in conjunction with a 50 Mc. third -overtone crystal connected between collector and base of the drift transistor. Operating bias level is adjusted by a variable potentiometer. The low impedance base of the 2N384 amplifier is tapped on the oscillator coil to achieve a match to the higher impedance collector circuit of the oscillator. The amplifier collector output circuit is inductively coupled to the antenna. It may be seen that this configuration bears a close similarity to a vacuum tube circuit in that the emitter of the transistor resembles the cathode of the tube. The base may be compared to the grid, and the collector to the plate. A two stage modulator section provides sufficient gain to operate a dynamic micro- www.americanradiohistory.com HANDBOOK 50 Mc. Transmitter phone. The audio stages are tranformer coupled and base driven. A 1N34 diode is used as a high level positive peak loading device to prevent peak clipping at high modulation levels. Positive peak clipping is employed since the collector supply voltage is negative with respect to ground. A simple metering system permits the operator to moniFigure 3. REAR VIEW OF TRANSISTORIZED TRANSMITTER. The two r.f. transistors are mounted in sockets on L- shaped bracket at the center of the chassis. Directly below them is the oscillator bias -potentiometer. Across the rear edge of the chassis are the audio stages, with the power terminals on the rear apron of the chassis. Relative size of transmitter and com- ponents may be judged from comparison with standard coaxial receptacle at left of chassis. Oscillator stage is at right, with amplifier at left. 579 tor the collector current of the r.f. stages. The positive terminal of the power supply is at "ground," or chassis potential. If NPNtype transistors are substituted for the specified units, battery polarity must be reversed. Transmitter Construction The complete transmitter is built upon a small aluminum chassis measuring x 31/2" x 1" in size. The front panel measures 6" x 4 ". The two r.f, transistor sockets and ri. components are mounted on an L- shaped aluminum bracket centered on the chassis, measuring 2" high by 21/4" long. The right -angle portion of the bracket holding the crystal socket is 11/2" high by 1" wide. Miniature transistor sockets are mounted in the top corners of the bracket, with the oscillator bias control centered beneath them (figure 3) . www.americanradiohistory.com 580 THE RADIO Low Power Transmitters The transistorized audio section is placed across the rear of the chassis. Transformer leads pass through small rubber grommets to the under -chassis area. At one end of the chassis is an aluminum bracket holding the coaxial antenna receptacle. Small components are mounted under the chassis on phenolic terminal strips. Transmitter wiring is straightforward, and is done with #22 insulated wire. Coil data is given in figure 2. Shown in figures 5 and 6 is a simple voltage regulated power supply that provides 18 volts at 100 milliamperes. A 2N561 power transistor is used as a series regulator, with a 2N44 serving as a regulator driver stage. The control element is a Zener diode delivering a constant source of 14.7 volts, which is used to set the output voltage. As the transmitter is operating near maximum transistor voltage values, it is important that the power supply Figure 4. UNDER -CHASSIS VIEW OF TRANSMITTER. Miniature components are mounted on phenolic terminal strips beneath the chassis. "Clipping" diode is at right, behind slide switch. Audio leads are run in shielded wire. voltage remain constant under varying loads. A voltage surge could possibly damage the transistors in the transmitter at this relatively high operating potential. The power supply is built upon an aluminum chassis measuring 51/2" x 31/2" x 1 ". The 2N561 power transistor must be insulated from the chassis by means of mica shims or an anodized plate, as the collector element is bonded to the case of the unit. The power supply may be tested by placing a 350 ohm, 10 watt resistor across the output. 18 volts should be developed across the resistor. Transmitter Adjustment and Tune -up When the transmitter wiring is completed, it should be carefully checked, especially in the area of the transistor sockets. Insert the r.f. transistors and crystal in their sockets and turn the oscillator bias potentiometer to maximum resistance. Place the meter switch in the oscillator position. Use a 52 ohm, 1 -watt composition resistor across the antenna receptacle as a dummy load for these tests. Turn the transmitter on and adjust the oscillator tuning capacitor for oscillation ( jump in collector current) as noted on the meter. Ad- www.americanradiohistory.com HANDBOOK TI SRI 200 -Watt Transmitter - Figure S. SCHEMATIC, VOLTAGE REGULATED POWER SUPPLY. B -115 volt neon lamp in holder 5R, Silicon rectifier, 400 v. p.i.v., S00 ma. Sarkes -Tarzian CM -500 -4- T,- Filament transformer. 26.8 volts at 1 a. Triad F -40X Z, -Zener diode, 15 volts, 1/2 watts, Motorola 1.5M15Z (10% tolerance) just the bias potentiometer for about 5 milliamperes oscillator current. Now, place the meter switch in the amplifier position and adjust the oscillator tuning capacitor for maximum meter reading. Adjust the amplifier tuning capacitor for a meter dip. Finally, adjust the antenna loading until the meter indicates about 6 milliamperes, re- resonating the circuit with the collector tuning capacitor. A field strength meter is helpful for the initial tune -up. The signal may now be monitored in a nearby 50 Mc. receiver. Connect a dynamic microphone and modulate the transmitter, adjusting the audio gain control for good modulation. The transmitter is now ready to be connected to your station antenna. 28 -2 581 styling, this deluxe unit is designed around the 7270 beam power tube and is capable of a conservative input of 200 watts on phone, and 250 watts on c.w. The transmitter covers all amateur bands between 10 and 80 meters, is v.f.o. controlled, and incorporates speech clipping for maximum audio "punch." A semiconductor high voltage rectifier is used to reduce heat and to provide improved voltage regulation. "Break -in" c.w. keying is incorporated employing a time differential system that results in chirp-free, clickless keying. Band changing is simplified by ganging the exciter switching circuits with the final amplifier pinetwork so that single control adjustment is achieved. In short, the transmitter incorporates all modern techniques to make it an up -todate, valuable item of station equipment that will not become obsolete. Circuit Description A block diagram of the table top transmitter is shown in figure 8. Thirteen tubes are five in the r.f. section, five employed, in the audio section, and the remainder in the control and power supply section. A complete schematic is shown in figures 9 and 10. The RCA 7270 beam power tube is employed in the final amplifier stage. This compact tube has high -perveance and good power gain. It can be operated at full input above 50 Mc., and has a maximum plate dissipation of 90 watts. At a plate potential of 1000 volts, this miniature "bottle" is capable of 250 watts input on c.w., and 200 watts input on a.m. phone. In addition, the tube has triple base -pin connections for the screen grid to permit good r.f. grounding and has large plate radiating fins for effective cooling. The A Deluxe 200 -Watt Tabletop Transmitter This self contained, TVI- proof, tabletop transmitter is designed for the amateur who desires a compact station capable of running sufficient power to provide consistent results in today's busy amateur bands. Modern in Figure 6. VOLTAGE REGULATED POWER SUPPLY. The silicon rectifiers are mounted above the chassis for proper ventilation, with the two transistors directly in front. 2N561 transistor is insulated from the chassis by a mica shim. www.americanradiohistory.com 582 T H Low Power Transmitters compact size makes it especially effective in the high frequency portions of the communication spectrum. Driving requirements are modest and permit the use of a simple band switching exciter. The Exciter Section. The high stability, all band v.f.o. consists of a 6AH6 (V1) in a "hot cathode" circuit, followed by a 6CL6 (V2) crystal oscillator- buffer stage. Very high -C is used in the oscillator stage to swamp out variations and changes in stray circuit and tube capacitance. The frequency determining circuit operates on 80 meters (L1 and associated components) for 80, 40 and 10 meter transmitter operation, and on 40 meters (L_ and associated components) for 20 and 15 meter transmitter operation. The circuit is a modified version of the Clapp oscillator. The tuning rate for each amateur band is changed automatically so that each band is spread over the entire portion of the tuning dial. Use of the exceptionally smooth Eddystone dial with a turn indicator makes it possible to read the transmitter frequency within a kilocycle or two. The oscillator is keyed by a section of the 12ÁU7 keyer tube (V6) for c.w. operation. The crystal oscillator- buffer stage (6CL6, V_) employs a broadly tuned 7 Mc. plate circuit for operation on 40 meters and all E R A D I O higher bands. For 80 meter operation, switch section SIB inserts an r.f. choke in series with the tuned circuit, dropping the r.f. output on this band to the correct value, and eliminating the necessity of tracking the stage across the relatively wide band. Switch S2 disables the v.f.o. and converts tube Vo into a 3.5 Mc. crystal oscillator, with the choice of two crystal frequencies. Figure 7. MODERN 200 WATT ALL BAND TABLE -TOP TRANSMITTER. Complete TVI -proof phone and c.w. transmitter is housed in modern -style tabletop cabinet. V.f.o. controlled, the transmitter covers all amateur bands between 80 to 10 meters. High level plate modulation with speech clipping is used fcr ptione, and a time -sequence break -in keyer is featured for c.w. operation. A standard 10 /s" x 19" panel is used in cose rock mounting is desired. Multi -meter on the left reads grid and screen current of amplifier stage, or modulator plate current. Selector switch is at left, directly below main tuning dial. Controls across bottom of panel are (I to r.): audio -gain, microphone receptacle, filament on switch, amplifier plate tuning (top) and amplifier plate loading (bottom), bandswitch, v.f.o.- crystal switch (top) and amplifier arid tuning (bottom), power switch (Ss) and pilot light, c.w.- tune-a.m. switch, and key jack. Below the tuning dial to the right is the grid drive control, and at the for right is the plate meter, M2. www.americanradiohistory.com HANDBOOK VI "FO 200 -Watt Transmitter V2 OSC.-SI/F. Sz ® 7 V3 V VS MUL r. DR/VER RF AMP. ® ® 583 PI NET. Mf YER V7 VR VI 1 MOD. Ve SPEECH VP CLIPPER V10 OR/ VER 3 RC MIC FILTER GA IN CL/P. V12 V13 +400 V. -13PV. V 5 +1000 V. 1 i RIAS SUPPLY L V. SUPPLY 41.1 N.V. SUPPLY ' 115V 1 BLOCK DIAGRAM OF Figure 8. 200 WATT TABLE -TOP TRANSMITTER. The plate circuit of the 6CL6 multiplier stage (V3) is untuned for 80 and 40 meter operation, and is resonated to 14 Mc. for 20 and 10 meter operation by coil L4, and to 21 Mc. for 15 meter operation by coil L5. This stage is block -grid keyed for c.w. operation. A 2E26 (V4) is used as a driver for the 7270 amplifier. This stage is neutralized and operates "straight through" on all bands except 10 meters, where it acts as a doubler from 14 Mc. A potentiometer control (grid drive) in the screen circuit of the 2E26 determines the excitation level to the final amplifier stage. The 7270 (V3) serves as a neutralized amplifier on all bands. Grid, screen and plate current are monitored for proper operation. A pi- network output circuit permits operation into unbalanced loads having impedances in the range of 50 to 75 ohms, and an s.w.r. value of 2.5 to 1, or less. The screen circuit is protected by relay RY3 which is energized by application of primary power to the high voltage plate supply. Thus, screen voltage cannot be applied to the tube unless plate voltage is also applied. The Mode Switch, S3. For tune -up purposes, amplifier screen voltage is dropped to a low value by the c.w.- tune -a.m. mode switch section S3,,. In the c.w. position, protective cut -off bias is applied to the 7270 by switch section S{i,. For phone operation, the amplifier screen circuit is "self- modulated" by choke CHI placed in the circuit by switch section Sac. The keyer tube (V11, 12AU7) keys the oscillator in addition to the 6CL6 multiplier stage, and optimum break -in characteristics may be set by the variable potentiometer www.americanradiohistory.com $ O ¡-ó 2 wá ÁI=i ó a; é " C Cf 000 . 8 U. « '24 _p Ó p V ^ o Y > Y ,¡ V > « O l'.-Ci a, E'ó = 0, O o. i¡ Z C ó > O.WV »il l: ói e O m ° M M ck M J pyll V) ; nM 5 V, 0Y Ñ o.; É I, 1 O Ó O N : r 0 ..L ó °c ^ , e: G il N ^ O Y Y; ° «o^' :".ca E _n ° E~Qx>, òoÑ On r È Y ó w e7 4 0 Y 4 á .7 .ì ó Ì H ° ; áúZZ.n Ñ o r >Nr 73: i i 4i; ú°'>tc aaaa aa I I a a J 01 ' i C IC V + ^O N Ç to x KM V 0 ~ M a n G M O ry^ ^-vim, Ó O vM uó v, C > 7 p ... xó_ gE iÓ;: úóó N > r.....- . ú h aa^>G ó.C>. do 61....S. o.°j o El..' ñE.o3 E.ñ _ ' N = oa Nió anM.. 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N :ó Et .:áÑ =>E:.+qu o< )4... i" N( UdácE^Y°á ° ^3¢íiia°C CVC>i nao iNX«+eN v Y Sz_ n öñ,,,"'óQev ó3áQ É o I N I U N N N J >< j 44 2Z á E^m a CO á t O4 C ^ , ~ t it ° á =cC oiE ó«á« gÉ VYi N Y d^ .12 el N X www.americanradiohistory.com I. I. I. I I. m THE RADIO Low Power Transmitters 586 labelled adjust keying. Switch section S3A shorts out the keyer in the tune and a.m. modes. Switch sections StD and S3F disable the modulator and speech amplifier in the c.w. and tune positions. Switch section S3E activates the 400 volt power supply in the Figure 11. TOP VIEW OF 200 WATT TRANSMITTER. Layout of above-chassis ports is shown in this view. Amplifier compartment is center rear, with buffer compartment between it and v.f.o. box near panel. Main power transformer is to the left of amplifier compartment, with modulation transformer and 811 -A tubes to the right. Adjacent to the horizontal oscillator tube (on left of v.f.o. box) are the two 6CL6 amplifier stages. The low- voltage power supply components are on the separate chassis at the left. On the side of the main chassis is the large cutout for air intake to the ventilating fan. To the right of the v.f.o. box are located the 12AU7 keyer tube and the OB2 voltage regulator. The adjust "keying" and "clipping" controls are between these tubes and the shielded speech amplifier tubes on the far edge of the chassis. The 6DE7 is next to the modulator tubes. tune positions, and the supply is activated by push -to -talk relay RY1 when switch S3 is set in the phone position. In addition, the transmitter may also be activated by the panel mounted power switch, S5 which completes the relay control circuit. c.w. and The Modulator Section. A pair of zero bias 811A tubes (V11, V12) are used as class B modulators, eliminating the need of bias and screen power supplies which are costly and expensive. A dual -purpose 6DE7 (V10) serves as a speech amplifier and driver stage. A 6AL5 double diode (V9) is a low level audio peak clipper which serves to increase the average level of modulation. This stage is followed by a home-made low -pass filter that restricts all audio frequencies above 3000 cycles. A 12AX7 dual triode (V8) provides sufficient gain for proper transmitter operation from a low level crystal microphone. In addition, a microphone push -to -talk circuit may be used to energize d.c. relay RY1 by means of a microphone control switch. www.americanradiohistory.com HANDBOOK 200 -Watt Transmitter Power Supplies. A careful selection of power supply components makes it possible to build a transmitter of this capability in such a small space. A cooling fan has been incorporated to insure that proper movement of air is maintained, and components have been selected for adequate safety margins and cool operation. The chassis has several cut -out openings on the sides and top for air circulation, and the chassis bottom plate of the audio section is made of perforated aluminum. The low voltage and bias supplies are conventional; however, the high voltage supply makes use of a bridge circuit employing twelve miniature silicon diode rectifiers. The center tap of the transformer high voltage winding is not used, and the 5 -volt winding is employed only to light a panel indicator lamp when the high voltage is switched on. The high voltage rectifier "stack" is protected from accidental overloads by a 1/2- ampere fuse placed in the B -plus lead to the filter system. Three 40 pfd., 450 -volt electrolytic capacitors are placed in series to provide approximately 12 izfd. at a working voltage of 1350. The entire transmitter, including power supplies, is built upon a heavy aluminum chassis measuring 13" x 17" x 3" in size. Shielded, chassis-type construction is used, and no reliance is placed upon the cabinet for TVIreduction (figures 7 and 11) . The v.f.o. is Transmitter Construction built as a separate unit in a 3" x 4" x 5" aluminum box which is bolted to the main chassis behind the geared dial. The low voltage and bias supplies are built on an aluminum chassis measuring 4" x 7" x 11/2", and may be seen in figure 11. The 5V4 -GA rectifier tube (V13) is mounted "outboard" on a small L- shaped bracket beside the power transformer (T2), fitting in nicely between the supply chassis and the buffer stage. The supply leads are brought through grommeted holes in the main chassis to terminal strips placed on the side apron of the chassis. The 7270 amplifier stage is entirely enclosed in an aluminum box measuring 61/2 inches square and 614 inches high (figure 12) . The top and back of this enclosure are fabricated from a single piece of perforated aluminum. The other three sides of the box are formed from an aluminum sheet, while the main chassis serves as the bottom of the enclosure. The 2E26 buffer stage is mounted between the v.f.o. enclosure and the final amplifier compartment. The buffer tube is placed in a horizontal position to best isolate the input and output circuits, and to obtain short leads in the plate circuit. The enclosure has screened sides and measures 3" x 2" x 1/2 " in size. The 2E26 tube projects into the box, with the base connections remaining outside the box in close proximity to the neutralizing Figure 12. CLOSEUP OF FINAL The AMPLIFIER ASSEMBLY. top and front of the final amplifier enclosure been removed to show placement of major components. The tank coil is mounted in a ver- have tical position, bolted to the side wall of the box. The output loading capacitor is just below the section. capacitor is mounted on inbetween pillars sulated the 7270 tube and the tank tuning capacitor. Plate leads are made of 10 -meter The coil neutralizing silver -plated 587 copper strap. The perforated shield at front of photo covers the horizontally 2E26 buffer mounted tube. www.americanradiohistory.com www.americanradiohistory.com 200 -Watt Transmitter capacitor. The plate coils of the 2E26 are beneath the chassis, grouped about bandswitch section S11). The buffer tuning capacitor ( labelled grid tuning) is adjacent to the bandswitch ( figure 13) . Placement of the major components may be seen in figure 11. The audio section is on the right of the chassis (viewed from the rear) and is separated from the r.f. section by a partition running the entire depth of the chassis on the underside. The 0A2 voltage regulator tube (V7) and the 12AU7 differential keyer tube (V6) are also in the audio section. The remainder of the smaller components are mounted beneath the chassis ( figure 13 ). The modulator section is to the left, while along the opposite side of the chassis are located the small blower fan, the high voltage silicon rectifiers, and the large filter choke. The center portion of the chassis is reserved for the r.f. section of the transmitter. The 7270 socket is centered towards the rear, directly behind a horizontal partition that separates the final amplifier components from the exciter stages. Vent holes are cut in the side aprons of the chassis (figure 11) and are covered with screening. In order to mount the 6.3 volt filament transformer (T4) on the side apron, a hole is drilled in the side of the case and the transformer leads are brought out through this hole, instead of via the bottom hole. This same technique is used to mount the high voltage filter choke, CH_. To facilitate mounting these components, 6 -32 nuts are soldered to the mounting flanges to accept the mounting bolts. The panel layout is dictared by placement of the major components. The v.f.o. tuning dial is centered on the panel near the top to allow proper clearance for the drive mechanism. The dial drive shaft, therefore, determines the position of the dual v.f.o. tuning capacitor which is mounted inside the enclosed oscillator assembly. A flexible coupling is used to join the dial to the capacitor to provide proper shaft alignment and smooth tuning. The v.f.o. itself is built as a separate unit after the position of the oscillator tuning capacitor has been determined. Panel Layout and Bandswitch Placement 589 The power amplifier output loading capacitor, plate tuning capacitor, and pi- network coil switch (S1E) are controlled from the front panel by means of right -angle drive systems placed beneath the chassis. The bandswitch S1 (centered on the panel) drives the v.f.o. bandswitch through a right angle coupler, in addition to driving the pi- network switch of the amplifier stage. Two small bevel gears are used for the oscillator drive, one mounted on the main bandswitch assembly between segments SIB and SID, and the other placed on the shaft of switch SIA which is located in the v.f.o. compartment (figure 14) . The oscillator bandswitch is placed directly below the v.f.o. tuning capacitor, with its shaft on the same vertical center line as that of the capacitor. The switch projects down through a 3A-inch matching hole in the chassis, placing the shaft at right angles to the center line of the main bandswitch where it is driven by the bevel gears. The main bandswitch assembly passes along the center line of the chassis to the final amplifier area, extending through a shield partition which isolates the multiplier and driver coils. An added section of shaft coupling drives a set of Boston gears mounted on a small support bracket at the back of the chassis. The gears have a 1:2 step -down ratio, as the final amplifier bandswitch has 60- degree indexing, whereas the main bandswitch has 30- degree indexing. It is a good idea to assemble the chassis, panel, and v.f.o. box, and lay out the various gear drive systems before other holes are drilled or components mounted in place. The final amplifier tuning and loading capacitors are mounted alongside the pi- network coil and their shafts project into the under -chassis area where they are joined to right -angle drives which bring the controls to the front panel. The amplifier tuning capacitor is driven with a set of Boston gears having a 2:1 step up ratio so that the dial turns 360- degrees while the capacitor rotates 180- degrees. This makes for easier adjustment of the circuit. Placement of the remaining panel controls and meters is not critical and is dictated by good symmetry and eye -appeal. Panel and chassis should be drilled together so that all shaft holes are in alignment. Panel and chassis are held together by two 13 -inch aluminum angle brackets placed at the ends of the chassis. www.americanradiohistory.com 590 Low Power Transmitters THE RADIO Figure 14. INSIDE THE V.F.O. ENCLOSURE. The oscillator tube socket is mounted to the left wall of the box, with the tuning capacitor adjacent to the terminals. The two one-inch diameter ceramic coil forms are mounted to the opposite wall with the padding capacitors between them. At ftc bottom of the box is the oscillator bandswitch, driven from the main bondswitch below deck by right -angle gears. Extra bolts are used to fasten the sides of the box securely in place, and all paint is scraped off the mating areas to ensure good contact. www.americanradiohistory.com HANDBOOK 200 -Watt Transmitter 591 .0J.1 F 1 600V. L r-- EA. 7.001 47 K 1.0 JJF 600 l V. K7',4K [CH 5 1 i100 47K .01,600V. 1 K .OJJF 600 Figure 15. V. time to obtain a spacing of about 3/16 inch, leaving nine plates in all (4 rotor, 5 stator). The capacitor is connected to the low potential (pi- network) side of the plate blocking capacitor so that d.c. plate voltage does not appear across it. Oscillator Construction. The whole v.f.o. unit may be wired separate from the transmitter. The tuning capacitor is a dual 25 µpfd. unit, with two rotor plates removed from the front section which tunes the 40 meter coil (L0). Two ceramic coil forms are mounted on the wall of the v.f.o. box opposite the tuning capacitor and two MAPC-type adjustable padding capacitors are in a line between the coils. The oscillator tube socket is on the side wall below the tuning capacitor, and all associated resistors and capacitors are mounted at the socket, with the exception of the silver mica capacitors which make up the various tuned circuits. These are mounted on the band switch, or to the wires running between switch a SPEECH AMPLIFIER TERMINAL BOARD. Make of phenolic, or other insulating material. The transmitter is most easily worked upon if the heavy and Wiring transformers are left off the chassis until the very last. The v.f.o. and low voltage supplies can be wired and tested as separate units before they are affixed to the chassis. The socket for the 7270 tube is recessed so that the vent holes in the base are on the underside of the chassis for passage of air from the cooling fan. The variable neutralizing capacitor for the amplifier stage is mounted vertically between the socket and the plate tuning capacitor (figure 12) and is adjustable from beneath the chassis. Space is limited so a modified APC -type unit is used. It is a 50 µpfd. size, with plates removed two at Assembly 270 L_ J Transmitter ' www.americanradiohistory.com amplifier components (including the audio filter) are pre- assembled on a phenolic board which is mounted on the side apron of the chassis ( figure 15) . Small components are soldered directly to socket pins. Miniature transistor-type cathode bypass capacitors are used to conserve space. The clipping and keyer controls are mounted on the chassis deck, between the low level stages. Filter inductor (CH5) is made from a Stancor TA -27 audio transformer, using the entire secondary winding as the coil. The voltage dividers of the bias supply and the dropping resistor for the regulator tube are placed in this section. The Power Supply and Control Circuits. The 5+ CUT OUT (WARD 2-}"A 2" RECTIFIER BOARD FRAMEWORK TO N.V. 3 TO N.V. ALUMINUM NEAT BARRIER s 204 204 204 I. THE RADIO Low Power Transmitters 592 401r 450 V. 40Ur 4010 450V. 450 V. I 6-12 FILTER BOARD Figure 17. LAYOUT OF RECTIFIER AND FILTER BOARDS. and coils. All tuned circuit wiring is done with #14 solid tinned copper wire. The v.f.o. output lead to the next stage passes via a feed through insulator in the bottom of the box to the under -chassis area. Filament and power leads are brought out through a grommet to a terminal strip beneath the main chassis. The Exciter and Audio Circuits. The exciter wiring is straightforward. The slug-tuned exciter coils are grouped about the main band switch, and all r.f. leads are short and direct. All r.f. bypass capacitors are mounted directly on the socket terminals. Part of the speech twelve silicon diode rectifiers and the filter network are placed below the high voltage transformer. The diodes are mounted on a perforated frame made from a sheet of fiberglass or phenolic material (figure 17) . The diodes are supported by their leads from small, hollow rivets employed as connecting points. The diode leads should be left untrimmed, and the leads are grasped with a pliers during the soldering process to prevent the heat of the iron from injuring the diode. The diode mounting plate is attached to the side apron of the chassis in front of the large air vent, directly behind the ventilating fan. The main filter capacitor consists of three series connected 450 volt capacitors in parallel with three wirewound resistors. These components are wired as a unit and mounted on a phenolic board on the rear apron of the chassis, alongside the blower motor. The I/2ampere high voltage fuse holder is also mounted on this board. A small aluminum shield is placed between the resistors and capacitors to act as a heat barrier ( figure 17) Figure 16. COIL DATA. TABLE TOP TRANSMITTER. L1-40 turns #22 enameled wire on -17 turns #20 enameled wire on I" ceramic form I" ceramic form (National XR -62) Range: 3.5 -4.0 Mc. L2 (Notional XR -62) Range: 7.0 -7.175 Mc. L3-40 turns »28 enameled wire on (National XR -50) Ronge: 7.0 Mc. L4 -20 /2" form i form turns #20 enameled wire on /2" (National XR-50) Range: 14 Mc. -12 turns »20 enameled wire on /2" form (National XR -50) Range: 21 Mc. L6 -16 turns 3/4" diameter tapped at 9 and 12 turns from junction with LT ( #3011 B&W miniductor) L7-38 turns #24 tapped at center 1" diameter ( »3016 B&W miniductor) LB 851 B&W tank coil assembly L5 -» www.americanradiohistory.com HANDBOOK 200 -Watt Transmitter The a.c, line fuse holders, antenna connector, relay connector J3, and Hy-pass feedthrough capacitors for the power line are also mounted on the rear apron. The coaxial antenna relay (RY2) is placed on the outside of the apron with a right-angle fitting added so that the antenna connection is accessible when the transmitter is placed in its cabinet. Most of the power and control wiring follows along the front inside edge of the chassis. Shielded wire is used for the 7270 filament and screen leads, and filament circuits are wired with #14 wire. The screen capacitor of the 7270 stage consists of three separate .001 pfd., 3 kv. ceramic disc capacitors, one placed from each screen socket pin to ground. The multi -meter (M1) has a 5 milliampere movement, and is converted into a low range voltmeter by the addition of a 750 -ohm series resistor. The voltage drops across shunts placed in the grid and screen circuits of the amplifier, and the plate circuit of the class B modulator are measured in this fashion. The meter scale is 0 -10 milliamperes when switch S4 is in the grid position, 0 -40 milliamperes in the screen position, and 0 -400 milliamperes in the modulator position. Tuning and Adjusting the Transmitter When the transmitter is corn pleted, the wiring should be visually inspected, and circuits "rung out" with an ohmmeter. The next step is to test and adjust the v.f.o. The fuse should be left out of the primary circuit of the high voltage supply to disable this section and to ensure that relay RY3 remains open. The 2E26 screen control should be set to remove screen voltage. Starting with the 80 meter band and the v.f.o. dial set at the low end (maximum v.f.o. tuning capacitance), trimming capacitor C4 is adjusted for 3500 kc., as noted on a frequency meter. With the specified coils, the 80 meter band extends over the entire dial scale, with the slug almost out of the coil form. Once the coverage is set, the slug should be secured with an extra nut to prevent movement. Capacitor C4 should not be moved now, as it will be in the circuit for the 40 and 10 meter adjustments. Next, the bandswitch is placed in the 10 meter position, the dial set at the low frequency end, and trimmer capacitor C7 adjusted for 28.0 Mc., with the v.f.o. dial pointer in approximately the same position as for 3.5 Mc. The 10 meter 593 band will now extend over almost the entire scale. Next, the bandswitch is placed in the 40 meter position, and it will be noted that the 7.0 Mc. position will fall very near the 28 Mc. mark, with the 40 meter band spread over most of the scale. The bandswitch should now be placed on the 20 meter position, and trimmer capacitor Cs adjusted so that 14.0 Mc. falls near the 3.5 Mc. dial point. The 15 meter calibration is automatically set by this adjustment. Once the v.f.o. has been calibrated, the 80 meter exciter circuits are tuned by simply advancing the 2E26 screen voltage potentiometer and tuning the driver stage to resonance, as indicated by grid current of the 7270 tube. Grid current should be held to a maximum of 4 milliamperes. The bandswitch may now be set to 40 meters and buffer coil L3 adjusted for maximum amplifier grid current with the v.f.o. set at 7.15 Mc. The 14 Mc. adjustments are now made with the bandswitch in the 20 meter position, and coil L4 peaked for maximum amplifier grid current at 14.15 Mc. Finally, the bandswitch is set to 15 meters and the slug of coil L, is peaked at 21.2 Mc. The driver stage is, of course, resonated for each band. The ten meter band is tuned by merely peaking the driver stage. Check both the low and high ends of the 10 meter band and equalize the grid drive by slight adjustments to coils L3 and L4, The 2E26 stage is neutralized in the 20 meter position by placing a temporary grid meter in series with the "cold" end of the 22K grid resistor and adjusting the neutralizing capacitor for minimum meter kick when the plate circuit is tuned through resonance. Screen voltage should be removed for this test. This setting will hold for all bands. Grid current to the 2E26 should not run over 3 milliamperes. The amplifier stage is now neutralized in a similar manner, using meter M1 to observe action of the grid current. This adjustment should be done on the 10 meter band. The final amplifier should not be operated without a dummy antenna load of some kind. Two 100 -watt lamp bulbs in parallel at the end of a short length of coaxial line will make a satisfactory load for preliminary adjustment purposes. Place the high voltage primary fuse in its receptacle and set the function switch S3 to the tune position. Grid current will now be www.americanradiohistory.com THE RADIO Low Power Transmitters 594 NOTE: DIMENS IONS OF FLANGE TO FIT TUN /NG CAPACITOR TERMINALS. CAPACITORS BU /LT WITHIN E /MAC SOCKET. C - s I L I ANT. A + SCR. "PLATE" LINE Lx 5001 I IBIAS ,00 DRILL BOTH PLATES FOR INSULATED BOLTS AND BUSHING. D E %C. L SUB -CHA SS /S AREA. F ROUND CORNERS B+ STRIP LINE CAVITY LINE WITH FINGER STOCK ',CHOKE" LINE LI ANT DIMENSIONS A 10 ,Et/i11L'tB ByyCyyDyy The E Cy/D BMCI 2i 220 MC. 144 MC F 7} A E 21" RFC, F 4 - EQUIVALENT CIRCUIT Figure 18. SCHEMATIC AND EQUIVALENT CIRCUIT OF STRIP LINE AMPLIFIER. strip line amplifier is built within 3" x 5" x 13" aluminum chassis box (144 Mc.), or 2" x S" x 91/2" (220 Mc.). The plate tuning capacitor of the 144 Mc. assembly is a cut -down turn, Johnson 154 -11 having three plates, spaced 0.25" apart. The antenna "hairpin" loop is 4" long and 11/2" wide (144 Mc.), or 2" long 34" wide (220 Mc.) placed parallel to strip line. Antenna resonating capacitor C2 is 35 µold. for either amplifier. Plate choke RFC, is Ohmite 1 or Z -220. B -plus lead posses through insulated hole in chassis, or may pass through feed- through type capacitor for low voltage operation (500 volts or less). The screen bypass capacitors are built within the Eimac air system sockets. Input circuits and blower are placed in sub- chassis enclosure. Z -144 observed on the final stage. The power switch S5 is turned on energizing relay RY1, and the final amplifier resonated and loaded to a plate current of about 150 milliamperes. The series screen resistor used in the tune mode limits off - resonance amplifier plate current to less than 200 milliamperes. The screen voltage tap on the 2500 ohm, 25 watt resistor is now adjusted (with the transmitter off!) to place about 320 volts on the 7270 screen circuit with the function switch in the a.m. position, and the amplifier loaded to 200 milliamperes plate current. In the c.u'. position the screen voltage will be slightly higher. Maximum voltage (400 volts) is always applied to the plate of the 2E26, and the dropping resistors reduce this to about 260 volts for the v.f.o., 6CL6 stages, and speech amplifier. The final plate voltage runs 1000 volts at a load current of 200 ma., and rises to about 1200 volts in the c.w., key -up position. Oscillator screen voltage is regulated at 105 volts. The bias supply delivers -135 volts, and the push -to -talk relay circuit is tapped down on the bleed resistor to supply about 100 volts to the d.c. relay RY1. The c.w. keying characteristic is determined by the adjustment of the keyer potentiometer, and by the choice of the 0.1 pfd. capacitors in the grid returns of the keyed tubes. For break in keying the "key -up" signal is monitored in the receiver and the keyer potentiometer is backed off until the oscillator signal just disappears. For phone operation, the modulator resting plate current is about 20 ma., kicking up to approximately 175 ma. on voice peaks. Maximum current excursions and modulation level are set by the adjust clip control, and the degree of modulation by the audio control. www.americanradiohistory.com HANDBOOK 28 -3 Strip Line Circuit Strip -Line Amplifiers for VHF Circuits A major stumbling block in the design and construction of high power v.h.f. transmitting equipment is the assembly of a suitable amplifier plate tank circuit. Simple L -C tuned circuits tend to assume microscopic proportions in this region of the spectrum and are incapable of handling large amounts of r.f. energy. Coaxial circuits, on the other hand, work well but are expensive, difficult to build and bulky to handle. A welcome compromise design is the simple strip line tank circuit, illustrated in figure 18A. The circuit is a modified cavity, making use of an inexpensive aluminum chassis as the outer enclosure, and employing strips of aluminum as the plate inductance. The line assumes r.f. ground potential at the end opposite the tube and is an approximate electrical eighth -wavelength long. It becomes an electrical quarter -wavelength when loaded by the tube and tuning capacitor placed at the high impedance end of the line. The line is made of two aluminum plates, separated by insulating material. This "sandwich" may be visualized as the equivalent circuit of figure 18B, which permits plate voltage to be applied to the amplifier tube via "plate line" L1 yet isolates the tuning capacitor and plate inductance from the d.c. voltage by means of a "distributed" r.f. choke. The cavity is completed by placing an aluminum cover plate over the open side of the chassis. A proper ratio of strip length and width compared to the cavity dimensions must be observed to determine the optimum line impedance, but the parameters may be varied sufficiently to permit the use of an inexpensive ready -made chassis for the line cavity without appreciable circuit degradation. Efficiency of the strip line is high, comparing favorably with conventional tank circuits operating at intermediate frequencies. The approximate characteristic impedance of the strip line may be determined from the following formula: 377 S (1) Z "=" W where S is the spacing between the strip line and the chassis, and W is the width of the strip; the width being much greater than the 595 spacing. A practical strip line will be shorter than a quarter wavelength by virtue of the interelectrode capacitance loading of the associated tube and the auxiliary tuning capacitor placed across the line (figure 18A) . In this case, the characteristic impedance of a loaded strip line is approximately: (2) Z = Z0 tan 131 0 p = , ) is the wavelength in centimeters and 1 is the length of the line in centimeters. If the total capacitive reactance is set equal to Z0, then tan /3, = 1, when the line length is 1/2- wavelength. For example: Assume a 1/2- wavelength line having a width (W) of 3 inches and a spacing (S) of 1 inch. The impedance, Zo is therefore (by formula 1) about 127 ohms. The output capacitance of a 4X250B is approximately 5 µµfd., representing an impedance value of about 220 ohms at 144 Mc. A parallel tuning capacitance of 5 µµfd. has the same impedance value, and the combined parallel impedance is approximately 110 ohms. Therefore a 1/2- wavelength line of the aforementioned dimensions could be used to tune the 2 meter band with a 4X250B tube. This line would be about 10 inches long, so a standard chassis box measuring 3" x 5" x 13" could be used for the plate cavity assembly. The construction of such a unit is described in this section. where Building the Strip Line 3 Shown in figure 19 are two strip line units for 144 Mc. Circuit and 220 Mc. The amplifiers are designed around the ceramic 4CX250B tube and may be operated at power inputs up to 500 watts for c.w. service, or 300 watts for a.m. phone. The limiting factors for power input are the maximum voltage rating of the plate bypass capacitor (if used), tuning capacitor spacing, and the voltage breakdown of the material employed as the dielectric of the strip line circuit. The units illustrated employ 10 -mil teflon coated fiberglass as the strip line dielectric, with fiber or teflon bushings and 4 -40 machine screws holding the assembly together. It is also possible to purchase teflon screws which could be used to advantage in this assembly. A sheet of 10 -mil mylar may be substituted for the fiberglass. Layout of the strip line units is shown in www.americanradiohistory.com 596 THE RADIO Low Power Transmitters Figure 19. STRIP LINE AMPLIFIERS FOR 144 MC. AND 220 MC. The simple mechanical assembly of the strip line tank circuit is especially suitable for home construction. Using o standard aluminum chassis as the foundation, the strip line consists of two aluminum plates separated by a dielectric. The line is supported from one end of the chassis box, and the tube socket is mounted in the bottom, with the tuning capacitor at the opposite end. At the near end of the assembly are the antenna resonating capacitor, the B -plus terminal and the antenna coaxial receptacle. The tube plate "finger stock" connector is made by EitelMcCullough, Inc., San Carlos, California, part =008294, Anode Collet. The dielectric material for the "sandwich" may be either 10 -mil (0.01 ") Mylar sheet, or 10 -mll teflon coated fiberglass. The mylar may be obtained from: Milam Co., 1100 Elmwood St., Providence, R. I. The teflon coated fiberglass may be obtained from Dodge Fibers, Inc., Hoosick Falls, N. Y. For maximum values of plate voltages, two layers of material should be used. Open side of chassis is closed by cover plate. figure 18. The "plate" section of the line (L2) is bolted to one end of the chassis box, at the proper height to encircle the anode of the tube without actually touching it. The "hot" end of this line is affixed to the stator of the plate tuning capacitor. The capacitor of the 144 Mc. amplifier has 0.25" spacing, as the unit is designed for high power operation. The "choke" plate of the "sandwich" line (L1) is shorter in length and spaced away from the grounded plate by means of the sheet fiberglass or mylar insulator. One end of this plate is connected to the B- supply through an auxiliary r.f. choke, and the opposite end makes contact to the anode of the tube by means of flexible metal finger stock soldered to the plate (see parts list) Both plates are sanded smooth to ensure that no metallic splinters or grains can puncture the thin dielectric sheet. The 220 Mc. unit is designed for low power doubler service at 500 volts and therefore makes use of a receiving -type capacitor in the plate circuit. A capacitor having greater spacing would be required for high voltage operation. The strip line amplifiers employ standard Eimac v.h.f. air sockets to ensure stability of operation. A standard grid circuit is employed and if neutralization is desired, it is possible to insert a probe into the strip line cavity and . www.americanradiohistory.com HANDBOOK "9T0" Electronic feed back a small amount of energy in the proper phase to the grid circuit. A "hairpin" loop (L3) provides coupling to the antenna circuit, and the reactance of the loop is tuned out by means of a series capacitor. The grid circuit components are built within a small chassis box placed beneath the strip line assembly, with a cooling blower mounted on the side of the box. The dimensions given are correct for the 4X150A- 4CX250B type tube, but may be varied for other tubes having slightly different interelectrode capacitances. Length of the strip line and the value of the tuning capacitor determine the resonant frequency, with the width of the center line and chassis spacing determining line impedance and exhibiting a second order frequency effect. It is therefore possible to effect small changes in the frequency of the circuit by varying the value of the tuning capacitor or the width and chassis spacing of the line if it is mechanically awkward to adjust the length of the strip. 28-4 A "9T0" Electronic Key The good c.w. operator is always trying to improve his skill and increase his keying speed. The modern way to do this is to use an electronic key. The dots, dashes, and spaces are all created electronically with a minimum Key 597 of effort on the part of the operator. A good keyer has a "mechanical mind of its own" and almost teaches the operator to send good code! Shown in this section is a version of the famous "9T0" keyer which provides the ultimate in reliable, precise electronic code. The keyer uses four tubes and two voltage regulators, and is packaged in a cabinet only slightly larger than a mechanical "bug" key. Best of all, it is inexpensive to build and fool -proof in operation. Operation of the Keyer One of the most reliable and stable methods of generating automatic and self- completing dots and dashes is the multivibrator system used in this keyer (figure 21). The keyer is driven by a "sideswiper" key which completes a control circuit to ground in either the "dot" or "dash" position. Closing the key on the "dot" side energizes the dot keyer tube (VOA) which turns on the dot multivibrator tube (VIA_B) to form a string of evenly spaced dots. Once the action has started, this generator will continue to form dots as long as the key contact is closed and will complete a full dot even if the key is released in the middle of a dot or a space. The output of the dot multivibrator is fed to the grid of the relay tube (VIA), and the contacts of the quick- acting relay in the plate circuit are used Figure 20. THE "9T0" ELECTRONIC KEY. simple, inexpensive electronic key generates dots, dashes, and spaces with a minimum effort on the part of the operator. Four tubes and two voltage regulators are used in a simple and reThis liable circuit. The "side swiper" key is mounted to an extension of the bottom plate of the keyer, making a unit only slightly larger than "bug" key. Panel controls are (I. to a mechanical r.): Weight control (with on -off switch), monitor speaker and speed control. Below these are the or zero -beat, tune-up button, and the neon character indicator. www.americanradiohistory.com THE RADIO Low Power Transmitters 598 +150 22 0. VIA -12AU7 1,200 V. T2 SPAR V3B - 12AU7- V3A V1B V B+ HI - e 3 VA V4B .005 \-12AU7J ìF = MATCHED PAIR RESISTORS ALL RESISTORS V2A 11SV.1, 2200 21e .w SRI T1 S, WATT UNLESS OTHERWISE NOTED M+ (ON WEIGHT CONTROL) + 5os í50 V H 5,1 I B+ 750 v 0A2 2..7 0 5,1 oOLr OB2 SO V. 6.3V. FILAMENTS +N 2,4,7 108V SR2 Figure 21. SCHEMATIC, ELECTRONIC KEY. RYI -DPST, 5000 ohm relay. Potter -Brumfield SM -SLS. Other satisfactory (but larger) relays are: Claire HG-1002 or W. E. 2766. The 15K series resistor may have to be adjusted for different relay models. Weight of dots may be varied by changing value of this resistor. 3- SRI, Silicon rectifier. p.i.v. 400 volts @ S00 ma. Sackes-Tarzian «M -500. TI -150 v. @ 50 ma., 6.3 v. @ 2a. Stancor PA -8421 T2- Push -pull replacement output transformer. Stancor A -3856 Key- Autronic sideswipes. Electrophysics Corp., 2500 West Coast Highway, Beach, California to key the transmitter and to activate an audio tone oscillator (V4B) used as a monitor. When the key is closed in the "dash" position, the dash keyer tube (V.,B) is energized, placing the dash multivibrator tube (V ;i A.B ) in readiness for operation, and at the same time sending a pulse through the 1N34 diode to start the dot multivibrator circuit again. This, in turn, triggers the dash multivibrator, Newport turning it on with the start of the first dot pulse, and turning it off with the end of the second dot pulse. The dash multivibrator, therefore, is an electronic switch which is turned on and off by two dot pulses. A dash of proper length and timing is created in this manner because the time length of the second "dot" adds to the "on" time of the switch circuit in holding the relay closed for the dash. www.americanradiohistory.com HANDBOOK "9T0" Electronic Key 599 Keyer Construction and Wiring Figure 23. TERMINAL BOARD LAYOUT. The parts shown outside of board are mounted underneath it. The lines indicate connections made to the board from tube pins or other components. The complete keyer configuration makes use of four 12ÁU7 double triode tubes. The power supply uses two silicon diodes to furnish both a positive and a negative voltage, regulated by the 0A2 and OB2 regulator tubes. Unregulated voltage is supplied to the relay tube and the tone oscillator. If desired, the 5963 computer type tube may be substituted for Vt, V., and V3 for improved long term stability of operation. The electronic keyer is built upon an aluminum chassis measuring 6" x 4" x 2 ", having two auxiliary end plates 5 inches high. A wrapover perforated aluminum cover screens the top and sides providing maximum ventilation. In addition, four large holes are punched in the sides of the chassis for additional cooling. The "sideswiper" key is mounted on an extension of the bottom plate of the chassis. The wiring of the keyer is simplified by mounting most of the multivibrator components on a terminal board placed in the underchassis area ( figure 22) The board is mounted on two pillars in the front -center of the chassis after all other wiring has been done (figure 23). The balance control is mounted on the rear apron of the chassis, as it requires adjustment only at intervals as the tubes age. When the unit is completed, all wiring should be checked. The unit is turned on and after a short warm -up period the key lever is held in the dash position and the balance control adjusted until self -completing dashes are formed. The neon lamp will flash at the character rate. The speed and weight controls are adjusted to suit the individual taste of the operator. Figure 22. UN DE RCHASSIS VIEW OF KEYER. The resistors and capacitors making up the mul- tivibrators are mounted on a terminal board sup- +os, 004r 04. ported below the chassis on short pillars. The silicon diode power rectifiers are on the side apron of the chassis adjacent to the filter capacitors. The balance potentiometer is on the rear apron between the keying lead and the .F') i Z.Z.Z. !°C 9/274 www.americanradiohistory.com Nr k ___/ - -¡_ -I wtrMu t4 power cord. . 1 1 1 i: wAV I C -_ ' t.Y c 600 Low Power Transmitters Figure 24. TOP VIEW OF KEYER. The keyer is built upon a 4" x 6" aluminum chassis. Layout of parts is not crowded. The audio oscillator transformer is near the front panel below the controls, and the sealed high -speed relay is in the center of the chassis with the 12AU7 tubes on either side. The two regulator tubes are between the power transformer and the rear panel. www.americanradiohistory.com OOdFPrNtNd.Otnt"MSa00rOdMMOdOrÑÿ.OOÑMOOF.O. Q .DrONPtfJMO`OdNrOPSF.CUJtNddMMNNNNrr.r..wr00 MtnSNr0p000CMCINVJTtNCIrrNdFrtn0.000tNNOF.OdNrO'oF F OVJ AddMMNNNCI mE Q.ó T `'V m iL 6 dFt+fNM.OrFMOSO .CSdMdFrFMr00.OVfdMNNrrr00 %U<Ç FrOFOtAS0FPtffdNFNFMrS00dMNNrr000000000000 © Vm U y E t. - ndtAFNFMrv7vfdMNNr.+ dMNNr.... . . . . . . tf. G t-U OO ' dMTMVfSOVJF OPOMPNCOOFMdPC1NFdM00 NtnOtN00dOSrdFdP DrrNtN.O.OFOC100Fd rrNNMdt'J.DSON.DOtNNOrtNNMOdFrOCdrdC1FM0000d`O.+ O OÑ r rrNNMdNDK0MOOOMrNO7f01Fr.0uOMMNST rrrNNMd0000M0r0MNMF0.0 rrrNNMdtN.DSO . O.OPSPtnMM U U Q G ,4 I I I PCJMMd.OFP=OSFSOrdtNMNPFd00FFSFFP`OFN tAMPFOdOrNVfOMOMP.OMFOOON rr-.NNMdtNFPrdFCJVFrbrFSM rr NNdDOPOOr ..PN G ,...,-.0.,0 V O m Ci PPNPPnSDSS dMdSCdF .1,-,--..--,...,0.0... 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Cl) EF- O.O.OFOtnOOCPNOddrOMMFOPF.ON d O I I I I FPONMtn.SNm M N N M M d rNdN iii 0.-2 N ecciti O O O O O O O O O O d 0 S F F M S d S N O N P d O d r P.O O P M O P rtN PtH N P PFddOVJNrPSMMFOUfSdNSNrdOONtIfOVfNOFOtNMMNrrr 0M0FrC1S0OMN0rrCl0O.0NOL0MNNr... U ONC.00ÑdMÑÑ....M.CSOtOdMNNr-r 0-.- Á-22 Ui O4 CZ o 0 F MN OrMVJNOMNFS -dtllOdrtASFtAFNrFVJOSFd m 8 O r C00000 0M0-.0 .0SFN.00PO.OtnF00dOd.OMNVfS0r NOMd MOdMPOMtfJdPFSPOOSNMSPdMNrPPN`ONOPPOMOOdPNr Otn Ot9 S N N O F N d N r O S F FO N N d M M M P FPdr N d S d rO OrdF O P S FO N N d d M M N N N N N N N Ñ S.D d N rNMd001-00,0rNMdN.OFSP0rNMd001.-00,OrNM0,00F00,0 NNNNNNNNNNMMMMMMI+fMMMd pi www.americanradiohistory.com CHAPTER TWENTY -NINE The trend in design of transmitters for operation on the high frequency bands is toward the use of a single high -level stage. The most common and most flexible arrangement includes a compact bandswitching exciter unit, with 15 to 100 watts output on all the high frequency bands, followed by a single power amplifier stage. In many cases the exciter unit is placed upon the operating table, with a coaxial cable feeding the drive to the power amplifier, although some operators prefer to have the exciter unit included in the main transmitter housing. This trend is a natural outgrowth of the increasing importance of v -f -o operation on the amateur bands. It is not practical to make a quick change in the operating frequency of a transmitter when a whole succession of stages must be returned to resonance following the frequency change. Another significant factor in implementing the trend has been the wide acceptance of commercially produced 75 and 150 -watt transmitters. These units provide r-f excitation and audio driving power for high level amplifiers running up to the 1000 -watt power limit. The amplifiers shown in this chapter may be easily driven by such exciters. 29 -1 Power Amplifier Design Choice of Tubes Either tetrode or triode tubes may be used in high -frequency power amplifiers. The choice is usually dependent upon the amount of driving power that is available for the power amplifier. If a transmitter -exciter of 100 -watt power capability is at hand (such as the Heath TX-1) it would be wise to employ a power amplifier whose grid driving requirements fall in the same range as the output power of the exciter. Triode tubes running 1- kilowatt input (plate modulated) generally require some 50 to 80 watts of grid driving power. Such a requirement is easily met by the output level of the 100 -watt transmitter which should 602 www.americanradiohistory.com Design ANTENNA CIRCUIT TO ANTENNA TO CIRCUIT i t BALANCED TWIN- LINE ,FED SYSTEM UNBALANCED COAXIAL FEED SYSTEM Figure 1 LINK COUPLED OUTPUT CIRCUITS FOR 603 PUSH -PULL AMPLIFIERS be employed as the exciter. Tetrode tubes (such as the 4 -250A) require only 10 to 15 watts of actual drive from the exciter for proper operation of the amplifier stage at 1kilowatt input. This means that the output from the 100 -watt transmitter has to be cut down to the 15 watt driving level. This is a nuisance, as it requires the addition of swamping resistors to the output circuit of the transmitter-exciter. The triode tubes, therefore, would lend themselves to a much more convenient driving arrangement than would the tetrode tubes, simply because their grid drive requirements fall within the power output range of the exciter unit. On the other hand, if the transmitter- exciter output level is of the order of 15 - 40 watts (the Johnson Ranger, for example) sufficient drive for triode tubes running 1- kilowatt input would be lacking. Tetrode tubes requiring low grid driving power would have to be employed in a high -level stage, or smaller triode tubes requiring modest grid drive and running 250 watts or so would have to be used. Power Amplifier Either push -pull or single ended circuits may be emof Circuits ployed in the power amplifier. Using modern tubes and properly designed circuits, either type is capable of high efficiency operation and low harmonic output. Push -pull circuits, whether using triode or tetrode tubes usually employ link coupling between the amplifier stage and the feed line running to the antenna or the antenna tuner. It is possible to use the link circuit in either an unbalanced or balanced configuration, as shown in figure 1, using unbalanced coaxial line, or balanced twin -line. Design- C,oice Figure 2 CONVENTIONAL PUSH -PULL AMPLIFIER CIRCUIT The mechanical layout should be symmetrical and the output coupling provision must be evenly balanced with respect to the plate coil C:- Approx. 1.5 1I.Id. per meter of wavelength per section Cr -Refer to plate tank capacitor design in Chapter 11 Cr-May be S00 111.ld., 10,000 -volt type ceramic capacitor NC -Max. usable capacitance should be greater, and min. capacitance less than rated grid -plate capacity of tubes in amplifier. 50,-, greater air gap than C,. R,-100 ohms, 20 watts. This resistor serves as low Q r -f choke. RFC, -band r -f choke suitable for plate current of tubes M, Suitable meters for d -c grid and plate currents All low voltage .001 pfd. and .01 pfd. by -pass capacitors are ceramic disc units (Centrolab DD or equiv.) L, -50 -watt plug -in coil, center link L-.- Plug -in coil, center link, of suitable power -All -M- rating. Common technique is to employ plug -in plate coils with the push -pull amplifier stage. This necessitates some kind of opening for coil changing purposes in the "electrically tight" enclosure surrounding the amplifier stage. Care must be used in the design and construction of the door for this opening or leakage of harmonics through the opening will result, with the attendant TVI problems. Single ended amplifiers may also employ link -coupled output devices, although the trend is to use pi- network circuits in conjunction with single ended tetrode stages. A tapped or otherwise variable tank coil may be used which is adjustable from the front panel, eliminating the necessity of plug -in coils and openings into the shielded enclosure of the amplifier. Pi- network circuits are becoming increasingly popular as coaxial feed systems are coming into use to couple the output circuits of transmitters directly to the antenna. www.americanradiohistory.com 604 H.F. Power Amplifiers 29 -2 Push -Pull Triode Amplifiers Figure 2 shows a basic push -pull triode amplifier circuit. While variations in the method of applying plate and filament voltages and bias are sometimes found, the basic circuit remains the same in all amplifiers. The amplifier filament transformer should be placed right on the amplifier chassis in close proximity to the tubes. Short filament leads are necessary to prevent excessive voltage drop in the connecting leads, and also to prevent r -f pickup in the filament circuit. Long filament leads can often induce instability in an otherwise stable amplifier circuit, especially if the leads Filament Supply are exposed to the radiated field of the plate circuit of the amplifier stage. The filament voltage should be the correct value specified by the tube manufacturer when measured at the tube sockets. A filament transformer having a tapped primary often will be found useful in adjusting the filament voltage. When there is a choice of having the filament voltage slightly higher or slightly lower than normal, the higher voltage is preferable. If the amplifier is to be overloaded, a filament voltage slightly higher than the rated value will give greater tube life. Filament bypass capacitors should be low internal inductance units of approximately .01 p,fd. A separate capacitor should be used for each socket terminal. Lower values of capacitance should be avoided to prevent spurious resonances in the internal filament structure of the tube. Use heavy, shielded filament leads for low voltage drop and maximum circuit isolation. The series plate voltage feed shown in figure 2 is the most satisfactory method for push -pull stages. This method of feed puts high voltage on the plate tank coil, but since the r -f voltage on the coil is in itself sufficient reason for protecting the coil from accidental bodily contact, no additional protective arrangements are made necesPlate Feed sary by the use of series feed. The insulation in the plate supply circuit should be adequate for the voltages encountered. In general, the insulation should be rated to withstand at least four times the maximum d-c plate voltage. For safety, the plate meter should be placed in the cathode return lead, since there is danger of voltage break- down between a metal panel and the meter T H E R A D I O movement at plate voltages much higher than one thousand. The recommended method of obtaining bias for c -w or plate modulated telephony is to use just sufficient fixed bias to protect the tubes in the event of excitation failure, and to obtain the rest by the voltage drop caused by flow of rectified grid current through a grid resistor. If desired, the bias supply may be omitted for telephony if an overload relay is incorporated in the plate circuit of the amplifier, the relay being adjusted to trip immediately when excitation is reGrid Bias moved from the stage. The grid resistor R. serves effectively as an r -f choke in the grid circuit because the impressed r -f voltage is low, and the Q of the resistor is poor. No r -f choke need be used in the grid bias return lead of the amplifier, other than those necessary for harmonic suppression. The bias supply may be built upon the amplifier chassis if care is taken to prevent r -f from finding its way into the supply. Ample shielding and lead filtering must be employed for sufficient isolation. The Grid Circuit As the power in the grid circuit is much lower than in the plate circuit, it is customary to use a close- spaced split-stator grid capacitor with sufficient capacitance for operation on the lowest frequency band. A physically small capacitor has a greater ratio of maximum to minimum capacitance, and it is possible to obtain a unit that will be satisfactory on all bands from 10 to 80 meters without the need for auxiliary padding capacitors. The rotor of the grid capacitor is grounded, simplifying mounting of the capacitor and providing circuit balance and electrical symmetry. Grounding the rotor also helps to retard v -h-f parasitics by by- passing them to ground in the grid circuit. The L/C ratio in the grid circuit should be fairly low, and care should be taken that circuit resonance is not reached with the grid capacitor at minimum capacitance. That is a direct invitation for instability and parasitic oscillations in the stage. The grid coil may be wound of no. 14 wire for driving powers of up to 100 watts. To restrict the field and thus aid in neutralizing, the grid coil should be physically no larger than absolutely necessary. Circuit Layout The most important consideration in constructing a push -pull amplifier is to maintain electrical symmetry on both sides of the balanced cir- www.americanradiohistory.com HANDBOOK Design 605 cuit. Of utmost importance in maintaining electrical balance is the control of stray capacitance between each side of the circuit and ground. Large masses of metal placed near one side of the grid or plate circuits can cause serious unbalance, especially at the higher frequencies, where the tank capacitance between one side of the tuned circuit and ground is often quite small in itself. Capacitive unbalance most often occurs when a plate or grid coil is located with one of its ends close to a metal panel. The solution to this difficulty is to mount the coil parallel to the panel to make the capacitance to ground equal from each end of the coil, or to place a grounded piece of metal opposite the "free" end of the coil to accomplish a capacity balance. Whenever possible, the grid and plate coils should be mounted at right angles to each other, and should be separated far enough apart to reduce coupling between them to a minimum. Coupling between the grid and plate coils will tend to make neutralization frequency sensitive, and it will be necessary to readjust the neutralizing capacitors of the stage when changing bands. All r -f leads should be made as short and direct as possible. The leads from the tube grids or plates should be connected directly to their respective tank capacitors, and the leads between the tank capacitors and coils should be as heavy as the wire that is used in the coils themselves. Plate and grid leads to the tubes may be made of flexible tinned braid or flat copper strip. Neutralizing leads should run directly to the tube grids and plates and should be separate from the grid and plate leads to the tank circuits. Having a portion of the plate or grid connections to their tank circuits serve as part of a neutralizing lead can often result in amplifier instability at certain operating frequencies. Excitation In general it may be stated Requirements that the overall power requirement for grid circuit excitation to a push -pull triode amplifier is approximately 10 per cent of the amount of the power output of the stage. Tetrodes require about 1 per cent to 3 percent excitation, referred to the power output of the stage. Excessive excitation to pentodes or tetrodes will often result in, reduced power output and efficiency. Push -Pull Symmetry is the secret of suc Amplifier cessful amplifier design. Shown Construction in figure 3 is the top view of a 350 watt push -pull all band LAYOUT Figure 3 350 -WATT PUSH -PULL TRIODE AMPLIFIER OF Two 811 -A tubes are employed in this circuit. Plate tuning capacitor is at left of chassis, with swinging -link type plug -in coil assembly mounted above it. Rotor of split -stator capacitor may be insulated from ground to increase voltage breakdown rating of capacitor. Note that pickup link is series -tuned to reduce circuit reactance. One corner of rotor plate of series capacitor is bent so that capacitor shorts itself out at maximum capacitance. Grid circuit coil and capacitor are at right. Center- linked plug -in coil is employed. Parasitic chokes are placed in grid leads adjacent to the tube sockets, and tube filaments are bypassed to ground with .01 ,.fd. ceramic capacitors. Complete area above the chassis is enclosed with perforated screen to reduce radiation of r.f, energy. amplifier employing 811 -A tubes. The circuit corresponds to that shown in figure 2 except that the 811 -A's are zero bias tubes. The bias terminals of the circuit are therefore jumpered together and no external bias supply is required at plate potentials less than 1300 volts. All r -f components are mounted above deck. The plate circuit tuning capacitor and swinging link tank coil are to the left, with the two disc -type neutralizing capacitors between the tank circuit and the tubes. At the right of the chassis is the grid tank circuit. Small parasitic chokes may be seen between the tube sockets and the grid circuit. Plate and grid meters are placed in the under-chassis area where they are shielded from the r -f field of the amplifier. Larger triode tubes such as the 810 and 8000 make excellent r -f amplifiers at the kilowatt level, but care must be taken in amplifier layout as the inter -electrode capacitance of these tubes is quite high. One rube and one neutralizing capacitor is placed on each side of the tank circuit (figures 4 and 5) to permit very short interconnecting leads. The relative position of the tubes and capacitors is trans- www.americanradiohistory.com 606 Figure 4 UNIQUE CHASSIS LAYOUT PERMITS SHORT LEADS IN KILOWATT AMPLIFIER Large size components required for high level amplifier often complicate amplifier layout. In this design, the plate tank capacitor sits astride small chassis running lengthwise on main chassis. Inductor is mounted to phenolic plate atop capacitor. Variable link is panel driven through right -angle gear drive. Plate circuit is grounded by safety arm when panel door is opened. Note that plate capacitor is mounted on four TV -type capacitors which serve to bypass unit, and also act as supports. A small parasitic choke is visible next to the grid terminal of the 810 tube. posed on each side of the chassis, as shown in the illustrations. The plate tank coil is mounted parallel to the front panel of the amplifier on a phenolic plate supported by the tuning capacitor which sits atop a small chassis -type box. The grid circuit tuning capacitor is located within this box, as seen in figure 6. An external bias supply is required for proper amplifier operation. Operating voltages may be determined from the instruction sheets for the particular tube to be employed. Whenever the amplifier enclosure requires panel door for coil changing access it is wise to place a power interlock on the door that will turn off the high voltage supply whenever the door is open! a THE RADIO H.F. Power Amplifiers Figure 5 LEFT -HAND VIEW OF KILOWATT AMPLIFIER OF FIGURE 4 Above shielded meter box is the protective "micro- switch" which opens the primary power circuit when the panel door is not closed. Tube sockets are recessed in the chassis so that top of tube socket shells are about 1,2-inch above chassis level. On right side of amplifier (facing it from the rear) the tube socket is nearest the panel, with the neutralizing capacitor behind it. On the opposite side, the capacitor is nearest the panel with the tube directly behind it. This layout transposition produces very short neutralizing leads, since connections may be made through the stator of plate tuning capacitor. 29 -3 Push -Pull Tetrode Amplifiers Tetrode tubes may be employed in push -pull amplifiers, although the modern trend is to parallel operation of these tubes. A typical circuit for push -pull operation is shown in figure 7. The remarks concerning the filament supply, plate feed, and grid bias in Section 29 -2 apply equally to tetrode stages. Because of the high circuit gain of the tetrode amplifier, extreme care must be taken to limit interstage feedback to an absolute minimum. Many amateurs have had bad luck with tetrode tubes and have been plagued with parasitics and spurious oscillations. It must be remembered with high gain tubes of this type www.americanradiohistory.com HANDBOOT( P -P Tetrode Amplifier 607 Figure 7 Figure 6 UNDER CHASSIS VIEW OF 1- KILOWATT TRIODE AMPLIFIER The grid circuit tuning capacitor and plate circuit r -f choke are contained in the below chassis enclosure formed by a small chassis mounted at right angles to the front panel. The bondswitch coil assembly for the grid circuit is mounted on two brackets above this cutout. A metal screen attached to the bottom of the amplifier completes the TVI -proof enclosure. that almost full output can be obtained with practically zero grid excitation. Any minute amount of energy fed back from the plate circuit to the grid circuit can cause instability or oscillation. Unless suitable precautions are incorporated in the electrical and mechanical design of the amplifier, this energy feedback will inevitably occur. Fortunately these precautions are simple. The grid and filament circuits must be isolated from the plate circuit. This is done by placing these circuits in an "electrically tight" box. All leads departing from this box are by -passed and filtered so that no r -f energy can pass along the leads into the box. This restricts the energy leakage path between the plate and grid circuits to the residual plate -to -grid capacity of the tetrode tubes. This capacity is of the order of 0.25 µpfd. per tube, and under normal conditions is sufficient to produce a highly regenerative condition in the amplifier. Whether or not the amplifier will actually break into oscillation is dependent upon circuit losses and residual lead inductance of the stage. Suffice to say that unless the tubes are actually neutralized a condition exists that will lead to circuit instability and oscillation under certain operating conditions. With luck, and a CONVENTIONAL PUSH -PULL TETRODE AMPLIFIER CIRCUIT Push -pull amplifier uses many of the same required by triode tubes (see figure 2). Screen supply is also required. Blower for filament seals of tubes. C. -Low internal inductance capacitor, .001 pfd., SKV. Centralab type 858S -1000. NC -See text and figure 8. PC- Parasitic choke. 50 ohm, 2 -watt composition resistor wound with 3 turns =12 e. components B- wire. Note: Strap multiple screen terminals together at socket with Je" copper ribbon. Attach PC to center of strap. heavily loaded plate circuit, one might be able to use an un- neutralized push -pull tetrode amplifier stage and suffer no ill effects from the residual grid -plate feedback of the tubes. In fact, a minute amount of external feedback in the power leads to the amplifier may just (by chance) cancel out the inherent feedback of the amplifier circuit. Such a condition, however, results in an amplifier that is not "reproduceable." There is no guarantee that a duplicate amplifier will perform in the same, stable manner. This is the one, great reason that many amateurs having built a tetrode amplifier that "looks just like the one in the book" find out to their sorrow that it does not "work like the one in the book." This borderline situation can easily be overcome by the simple process of neutralizing the high -gain tetrode tubes. Once this is done, and the amplifier is tested for parasitic oscillations (and the oscillations eliminated if they occur) the tetrode amplifier will perform in an excellent manner on all bands. In a word, it will be "reproduceable." www.americanradiohistory.com 608 1 THE RADIO H.F. Power Amplifiers - ,,, .ita.... .: 1 1 Figure 9 UNDER CHASSIS VIEW OF 4 -250A AMPLIFIER Figure 8 REAR VIEW OF PUSH PULL 4 -250A AMPLIFIER The neutralizing rods are mounted on ceramic feedthrough insulators adjacent to each tube socket. Low voltage power leads leave the grid circuit compartment via Hypass capacitors located on the lower left corner of the chassis. A screen plate covers the rear of the amplifier during operation. This plate was removed for the photograph. The bias supply for the amplifier is mounted at the front of the chassis between the two control shafts. A blower motor is mounted beneath each tube socket. A screened plate is placed on the bottom of the chassis to complete the under -chassis shielding. - As a summation, three requirements must be met for proper operation of tetrode tubes whether in a push-pull or parallel mode: 1. Complete isolation must be achieved between the grid and plate circuits. 2. The tubes must be neutralized. 3. The circuit must be parasitic -free. Amplifier Construction The push -pull tetrode ampli fier should be built around two "r -f tight" boxes for the grid and plate circuits. A typical layout that has proven very satisfactory is shown in figures 8 and 9. The amplifier is designed around a Barker & Williamson "butterfly" tuning capacitor. The 4 -250A tetrode tubes are mounted at the rear of the chassis on each side of the capacitor. The base shells of the tubes are grounded by spring clips, and short adjustable rods project up beside each tube to act as neutralizing capacitors. The leads to these rods are cross connected beneath the chassis and the rods provide a small value of capacitance to the plates of the tubes. This neutralization is necessary when the tube is operated with high power gain and high screen voltage. As the operating frequency of the tube is increased, the inductance of the internal screen support lead of the tube becomes an important part of the screen ground return circuit. At some critical frequency (about 45 Mc. for the 4 -250A tube) the screen lead inductance causes a series resonant condition and the tube is said to be "self- neutralized" at this frequency. Above this frequency the screen of the tetrode tube cannot be held at ground potential by the usual screen by -pass capacitors. With normal circuitry, the tetrode tube will have a tendency to self -oscillate somewhere in the 120 Mc. to 160 Mc. region. Low capacity tetrodes that can operate efficiently at such a high frequency are capable of generating robust parasitic oscillations in this region while the operator is vainly trying to get them operating at some lower frequency. The solution is to introduce enough loss in the circuit at the frequency of the parasitic so as to render oscillation impossible. This procedure has been followed in this amplifier. During a long series of experiments designed to stabilize large tetrode tubes, it was found that suppression circuits were most effective when inserted in the screen lead of the www.americanradiohistory.com HANDBOOK Pi- Network Amplifiers tetrode. The screen, it seemed, would have r -f potentials measuring into the thousands of volts upon it during a period of parasitic oscillation. By- passing the screen to ground with copper strap connections and multiple by-pass capacitors did little to decrease the amplitude of the oscillation. Excellent parasitic suppression was brought about by strapping the screen leads of the 4 -250A socket together (figure 7) and inserting a parasitic choke between the screen terminal of the socket and the screen by -pass capacitor. After this was done, a very minor tendency towards self -oscillation was noted at extremely high plate voltages. A small parasitic choke in each grid lead of the 4 -250A tubes eliminated this completely. The neutralizing rods are mounted upon two feedthrough insulators and cross -connected to the 4 -250A control grids beneath the chassis. These rods are threaded so that they may be run up and down the insulator bolt for neutralizing adjustment. Because of the compact size of many tetrodes it is necessary to cool the filament seals of the tube with a blast of air. A small blower can be mounted beneath the chassis to project cooling air directly at the socket of the tube as shown in figure 9. Inductive Tuning of Push -Pull Amplifiers The plate tank circuit of the push -pull amplifier must have a low impedance to ground at harmonic frequencies to provide adequate harmonic suppression. The usual split- stator tank capacitor, however, has an uncommonly high impedance in the VHF region wherein the interference -causing harmonics lie. A push -pull vacuum-type capacitor may be used as these units have very low internal inductance, but the cost of such a capac- 609 seen in figure 10. Two fixed vacuum capacitors are mounted vertically upon the chassis and the upper terminals are attached to the plates of the amplifier tubes by means of low impedance straps. Resonance is established by rotation of a shorted copper loop located within the amplifier tank coil. This loop is made of a %8" long section of copper water pipe, two inches in diameter. Approximate resonance is established by varying the spacing between the turns of the copper tubing tank coil. Inductive coupling is used between the tank coil and the antenna circuit in the usual manner. Sufficient range to enable the operator to cover a complete high frequency band may be had with this interesting tuning method. 29 -4 Tetrode PiNetwork Amplifiers The most popular amplifier today for both commercial and amateur use is the pi- network configuration shown in figure 11. This circuit is especially suited to tetrode tubes, although triode tubes may be used under certain circumstances. A common form of pi- network amplifier is shown in figure 11A. The pi circuit forms the matching system between the plate of the amplifier tube and the low impedance, unbalanced antenna circuit. The coil and input capacitor itor is quite high. A novel solution to this problem is to employ a split stator capacitor made up of two inexpensive fixed vacuum capacitors. Amplifier adjustment can then best be accomplished by inductive tuning of the plate tank coil as Figure 10 INDUCTIVE TUNING MAY BE EMPLOYED IN HIGH POWER AMPLIFIER Two fixed vacuum capacitors form split -stator capacitance, providing very low inductance ground path for plate circuit harmonics. Tuning is accomplished by means of shorted, single -turn link placed in center of tank coil. Shorted link is made from 3k -inch section cut from copper water pipe. Larger link outside of tank coil is antenna pick -up coil. www.americanradiohistory.com 610 H.F. THE RADIO Power Amplifiers LOW Z OUTPUT EXCITATION - BIA' 15V.1, Figure 11 TYPICAL PI- NETWORK CONFIGURATIONS circuit provides out -of -phase voltage for grid neutralization of tetrode tube. Rotary coil employed in plate circuit, with small, fixed auxiliary coil for 28 Mc. Multiple tuning grid tank TI covers 3.5 - 30 Mc. without switching. Tapped grid and plate inductors are used with "bridge type" neutralizing circuit for tetrode amplifier stage. Vacuum tuning capacitor is used in input section of pi- network. Untuned input circuit (resistance loaded) and plate inductor ganged with tuning capacitor comprise simple amplifier configuration. PCr, PC.-57 ohm, 2 watt composition resistor, wound with 3 turns o 18 c. wire. A -Split grid is BC- www.americanradiohistory.com HANDBOOK Pi- Network of the pi may be varied to tune the circuit over a 10 to 1 frequency range (usually 3.0 - 30 Mc.) . Operation over the 20 - 30 Mc. range takes place when the variable slider on coil L2 is adjusted to short this coil out of the circuit. Coil L. therefore comprises the tank inductance for the highest portion of the operating range. This coil has no taps or sliders and is constructed for the highest possible Q at the high frequency end of the range. The adjustable coil ( because of the variable tap and physical construction) usually has a lower Q than that of the fixed coil. The degree of loading is controlled by capacitors C. and G. The amount of circuit capacity required at this point is inversely proportional to the operating frequency and to the impedance of the antenna circuit. A loading capacitor range of 100 µµfd. to 2500 µµfd. is normally ample to cover the 3.5 - 30 Mc. range. The pi circuit is usually shunt-fed to remove the d.c. plate voltage from the coils and capacitors. The components are held at ground potential by completing the circuit ground through the choke RFC,. Great stress is placed upon the plate circuit choke RFC:. This component must be specially designed for this mode of operation, having low inter -turn capacity and no spurious internal resonances throughout the operating range of the amplifier. Parasitic suppression is accomplished by means of chokes PC -I and PC -2 in the screen and grid leads of the tetrode. Suitable values for these chokes are given in the parts list of figure 1L Effective parasitic suppression is dependent to a large degree upon the choice of screen bypass capacitor C2. This component must have extremely low inductance throughout the operating range of the amplifier and well up into the VHF parasitic range. The capacitor must have a voltage rating equal to at least twice the screen potential (four times the screen potential for plate modulation). There are practically no capacitors available that will perform this difficult task. One satisfactory solution is to allow the amplifier chassis to form one plate of the screen capacitor. A "sandwich" is built upon the chassis with a sheet of insulating material of high dielectric constant and a matching metal sheet which forms the screen side of the capacitance. A capacitor of this type has very low internal inductance but is very bulky and takes up valuable space beneath the chassis. One suitable capacitor for this position is the Centralab type 858S-1000, Amplifiers 611 BIAS SUPPLY SCREEN SUPPLY P= PLATE SUPPLY B = S OPERATE TUNE "COMMON MINUS" LEAD BY- Figure 12 GROUNDED SCREEN GRID CONFIGURATION PROVIDES HIGH ORDER OF ISOLATION IN TETRODE AMPLIFIER STAGE A- Typical amplifier circuit has cathode return at ground potential. All circuits return to cathode. -All circuits return to cathode, but ground point has been shifted to screen terminal of tube. Operation of the circuit remains the same, as potential differences between elements of the tube are the same as in circuit A. C- Practical grounded screen circuit. "Common minus" lead returns to negative of plate supply, which cannot be grounded. Switch S: removes screen voltage for tuneB up purposes. rated at 1000 µµfd. at 5000 volts. This compact ceramic capacitor has relatively low internal inductance and may be mounted to the chassis by a 6 -32 bolt. It is shown in various amplifiers described in this chapter. Further screen isolation may be provided by a shielded power lead, isolated from the screen by a .001 pfd. ceramic capacitor and a 100 ohm carbon resistor. Various forms of the basic pi- network amplifier are shown in figure 11. The A configuration employs the so- called "all- band" grid tank circuit and a rotary pi- network coil in the www.americanradiohistory.com 612 THE RADIO H.F. Power Amplifiers plate circuit. The B circuit uses coil switching in the grid circuit, bridge neutralization, and a tapped pi- network coil with a vacuum tuning capacitor. Figure 1 1C shows an interesting circuit that is becoming more popular for class ABI linear operation. A tetrode tube operating under class ABI conditions draws no grid current and requires no grid driving power. Only r -f voltage is required for proper operation. It is possible therefore to dispense with the usual tuned grid circuit and neutralizing capacitor and in their place employ a simple load resistor in the grid circuit across which the required excitation voltage may be developed. This resistor can be of the order of 50 - 300 ohms, depending upon circuit requirements. Considerable power must be dissipated in the resistor to develop sufficicnt grid swing, but driving power is often cheaper to obtain than the cost of the usual grid circuit components. In addition, the low impedance grid return removes the tendency towards instability that is so common to the circuits of figure 1IA and 11B. Neutralization is not required of the circuit of figure 11C, and in many cases parasitic suppression may be omitted. The price that must be paid is the additional excitation that is required to develop operating voltage across grid resistor R.. The pi- network circuit of figure 11C is interesting in that the rotary coil L. and the plate tuning capacitor Ca are ganged together by a gear train, enabling the circuit to be tuned to resonance with one panel control instead of the two required by the circuit of figure 11A. Careful design of the rotary inductor will permit the elimination of the auxiliary high frequency coil L,, reducing the cost and complexity of the circuit. result of this loss of circuit isolation. A solution to this problem is to eliminate the screen bypass capacitor, grounding the screen terminals of the tube by means of a low inductance strap. Screen voltage is then applied to the tube by grounding the positive terminal of the screen supply, and "floating" the negative of the screen and bias supplies below ground potential as shown in figure 12. Meters are placed in the separate circuit cathode return leads, and each meter reads only the current flowing in that particular circuit. Operation of this grounded screen circuit is normal in all respects, and it may be applied to any form of grid- driven tetrode amplifier with good results. 29 -5 Grounded -Grid Amplifier Design The grounded grid (g -g) amplifier has achieved astounding popularity in recent years as a high power linear stage for sideband application. Various versions of this circuit are illustrated in figure 13. In the basic circuit, the control grid of the tube is at r.f. ground potential and the exciting signal is applied to the cathode by means of a tuned circuit. Since the grid of the tube is grounded, it serves as a shield between the input and output circuits, making neutralization unnecessary in many instances. The very small plate to cathode capacitance of most tubes permits a minimum of intrastage coupling below 30 Mc. In addition, when zero bias triodes or tetrodes are used, screen or bias supplies are not usually required. Configuration The Grounded Screen For maximum shielding, it is necessary to operate Feedthrough Power the tetrode tube with the screen at r.f. ground potential. As the screen has a d.c. potential applied to it (in grid driven circuits), it must be bypassed to ground to provide the necessary r.f. return. The bypass capacitor employed must perform efficiently over a vast frequency spectrum that includes the operating range, plus the region of possible v.h.f. parasitic oscillations. This is a large order, and the usual bypass capacitors possess sufficient inductance to introduce regeneration into the screen circuit, degrading the grid -plate shielding to a marked degree. Nonlinearity and self -oscillation can be the power appears in the plate circuit of the grounded grid (cathode driven) amplifier and is termed feedthrough power. In any amplifier of this type, whether it be triode or tetrode, it is desirable to have a large ratio of feedthrough power to peak grid driving power. The feedthrough power acts as a swamping resistor across the driving circuit to stabilize the effects of grid loading. The ratio of feedthrough power to driving power should be about 10 to 1 for best stage linearity. The feedthrough power provides the user with added output power he would not obtain from a more conventional circuit. The www.americanradiohistory.com A portion of the exciting HANDBOOK Grounded -Grid Amplifier R.r. OUT 613 R.r. OUT EXC E xC. R.r. OUT EXC. © Figure 13 THE GROUNDED-GRID AMPLIFIER linear amplifier for sideband service, the grounded -grid circuit provides economy Widely used as a and simplicity, in addition to a worthwhile reduction in intermodulation distortion. A -The basic g -g amplifier employs tuned input circuit. B simplified circuit employs untuned r.f. choke in cathode in place of the tuned circuit. Linearity and power output are inferior compared to circuit of figure A. Simple high -C pi- network may be used to match output impedance of sideband exciter to input impedance of grounded-grid stage. Parallel- tuned, High -C circuit may be employed for bandswitching amplifier. Excitation tap is adjusted to provide low value of sm.,. on exciter coaxial line. -A C- D- driver stage for the grounded grid amplifier must, of cous-se, supply the normal excitation power plus the feedthrough power. Many commercial sideband exciters have power output capabilities of the order of 70 to 100 watts and are thus well suited to drive high power grounded grid linear amplifier stages whose total excitation requirements fall within this range. Distortion Laboratory measurements made on various tubes in the circuit of figure 13A show that a distortion reduction of the order of 5 to 10 decibels in odd -order products can be obtained by operating the tube in grounded grid service as opposed to grid- driven service. The improvement in distortion varies from tube type to tube type, but some order of improvement is noted for all tube types tested. Most amateur -type transmitting tubes provide signal -to- distortion ratios of -20 to -30 decibels at full output in class ABI grid- driven operation. The ratio increases to approximately -25 to -40 deciProducts bels for class B grounded grid operation. Dis- tortion improvement is substantial, but not as great as might otherwise be assumed from the large amount of feedback inherent in the grounded grid circuit. A simplified version of the grounded grid amplifier is shown in figure 13B. This configuration utilizes an untuned input circuit, and is very popular as an inexpensive and simplified form of the more sophisticated circuit of figure 13A. It has inherent limitations, however, that should be recognized. In general, slightly less power output and efficiency is observed with the untuned cathode circuit, odd -order distortion products run 4 to 6 decibels higher, and the circuit is harder to drive and match to the exciter than is the tuned cathode circuit of figure 13A. For maximum linearity and optimum operation, a certain amount of "flywheel" effect is required in the cathode input that can only be supplied by a high -C tuned circuit of some form. Since the single ended class B grounded grid linear amplifier draws grid current on only one -half (or less) of the operating cycle, www.americanradiohistory.com 614 THE RADIO H.F. Power Amplifiers -BIAS R.F.OUT. EXC. Figure 14 TETRODE TUBES MAY BE USED IN GROUNDED-GRID AMPLIFIERS Tetrodo tube may be used in cathode driven configuration, with bias and screen voltages applied to elements which are at r.f. ground potential. B-Grid current of grounded -grid tube is easily monitored by R -C network which lifts grid above ground sufficiently to permit a millivoltmeter to indicate voltage drop across -ohm resistor. Meter is a 0 -1 d.c. milliameter in series with appropriate multiplier resistor. A- 1 the sideband exciter "sees" a low impedance load during this time, and a very high impedance load over the balance of the cycle. Linearity of the exciter is thereby affected and the distortion products of the exciter are enhanced. Thus, the driving signal is degraded in the cathode circuit of the grounded grid stage unless the unbalanced input impedance can be modified in some fashion. A high -C tuned circuit stores enough energy over the operating r.f. cycle so that the exciter "sees" a relatively constant load at all times. In addition, the tuned circuit may be tapped or otherwise adjusted so that the standing wave ratio on the coaxial line coupling the exciter to the amplifier is relatively low. This is a great advantage, particularly in the case of those exciters having fixed -ratio pi- network output circuits designed expressly for a 50 -ohm termination. Finally, it must be noted that removal of the tuned cathode circuit breaks the amplifier plate circuit return to the cathode, and r.f. plate current pulses must return to the cathode via the outer shield of the driver coaxial line and back via the center conductor! Extreme fluctuations in exciter loading, intermodulation distortion, and TVI can be noticed by changing the length of the cable between the exciter and the grounded grid amplifier when an untuned cathode input circuit is employed. Design features of the single ended and push -pull ampliConstruction fiers discussed previously apply equally well to the grounded grid stage. The g -g linear amplifier Grounded Grid Amplifier may have either configuration, although the majority of g -g stages are single- ended, as push -pull offers no distinct advantages and adds greatly to circuit complexity. The cathode circuit of the amplifier is resonated to the operating frequency by means of a high -C tank (figure 13A). Resonance is indicated by maximum grid current of the stage. A low value of s.w.r. on the driver coaxial line may be achieved by adjusting the tap on the tuned circuit, or by varying the capacitors of the pi- network (figure 13C). Correct adjustments will produce minimum s.w.r. and maximum amplifier grid current at the same settings. The cathode tank should have a Q of 2 or more. The cathode circuit should be completely shielded from the plate circuit. It is common practice to mount the cathode components in an "r.f. tight" box below the chassis of the amplifier, and to place the plate circuit components in a screened box above the chassis. The grid (or screen) circuit of the tube is operated at r.f. ground potential, or may have d.c. voltage applied to it to determine the operating parameters of the stage (figure 14A ). In either case, the r.f. path to ground must be short, and have extremely low inductance, otherwise the screening action of the element will be impaired. The grid (and screen) therefore, must be bypassed to ground over a frequency range that includes the operating spectrum as well as the region of possible v.h.f. parasitic oscillations. This is quite a large order. The inherent inductance of the usual bypass capacitor plus the length of www.americanradiohistory.com HANDBOOK Grounded -Grid Amplifier element lead within the tube is often sufficient to introduce enough regeneration into the circuit to degrade the linearity of the amplifier at high signal levels even though the instability is not great enough to cause parasitic oscillation. In addition, it is often desired to "unground" the grounded screen or grid sufficiently to permit a metering circuit to be inserted. One practical solution to these problems is to shunt the tube element to ground by means of a 1 -ohm composition resistor, bypassed with a .01 µfd. ceramic disc capacitor. The voltage drop caused by the flow of grid (or screen) current through the resistor can easily be measured by a milli -voltmeter whose scale is calibrated in terms of element current (figure 14B). The plate circuit of the grounded grid amplifier is conventional, and either pi- network or inductive coupling to the load may be used. There is some evidence to support the belief that intermodulation distortion products are reduced by employing plate circuit Q's somewhat higher than normally used in class -C amplifier design. A circuit Q of 15 or greater is thus recommended for grounded grid amplifier plate circuits. Tuning the Since the input and output circuits of the grounded grid Amplifier amplifier are in series, a certain proportion of driving power appears in the output circuit. If full excitation is applied to the stage and the output circuit is opened, or the plate voltage removed from the tube, practically all of the driving power will be dissipated by the grid of the tube. Overheating of this element will quickly occur under these circumstances, followed by damage to the tube. Full excitation should therefore never be applied to a grounded grid stage unless plate voltage is applied beforehand, and the stage is loaded to the antenna. For best linearity, the output circuit of the grounded grid stage should be overcoupled so that power output drops about 2- percent from maximum value. A simple output r.f. voltmeter is indispensable for proper circuit adjustment. Excessive grid current is a sign of antenna undercoupling, and overcoupling is indicated by a rapid drop in output power. Proper grounded grid stage operation can be Grounded Grid 615 determined by finding the optimum ratio between grid and plate current and by adjusting the drive level and loading to maintain this ratio. Many manufacturers now provide grounded grid operation data for their tubes, and the ratio of grid to plate current can be determined from the data for each particular tube. Not all tubes are suitable for grounded grid service. In addition, the signal- to -disdistortion ratio of the suitable tubes varies over a wide range. Some of the best g -g performers are the 811A, 813, 7094, 4 -125A, 4-250A, 4 -400A and 4- 1000A. In addition, the 3 -400Z and 3 -1000Z triodes are specifically designed for low distortion, grounded grid amplifier service. The older types 837 and 803 are used extensively for g -g operation but are not recommended because of poor signal -to- distortion ratios. Certain types of tetrodes, exemplified by the 4 -65A, 4X150A, 4CX300A and 4CX1000A should not be used as grounded grid amplifiers unless grid bias and screen voltage are applied to the elements of the tube (figure 14A). The internal structure of these tubes permits unusually high values of grid current to flow when true grounded grid circuitry is used, and the tube may be easily damaged by this mode of operation. The efficiency of a typical grounded grid amplifier runs between 55- and 65-percent, indicating that the tube employed should have plenty of plate dissipation. In general, the p.e.p. input in watts to a tube operating in grounded grid configuration can safely be about 2.5 to 3 times the rated plate dissipaChoice of Tubes for G -G Service tion. Because of the relatively low average -topeak power of the human voice it is tempting to push this ratio to a higher figure in order to obtain more output from a given tube. This action is unwise in that the odd -order distortion products rise rapidly when the tube is overloaded, and because no safety margin is left for tuning errors or circuit adjustments. Neutralization At some high frequency the shielding action of the grid of the g -g amplifier deteriorates. Neutralization may be necessary at higher frequencies either because of the presence of inductance between the active grid of the G -G Stage www.americanradiohistory.com 616 THE RADIO H.F. Power Amplifiers n. r. ouT. EXC. O Figure 15 NEUTRALIZING CIRCUITS FOR GROUNDED -GRID STAGES Neutralization of the g -g stage may be necessary at the higher frequencies. Energy fed back in proper phase from plate to cathode is used to neutralize the unwanted energy fed through the tube (A). Reactance placed in series with the grid return lead (B) will accomplish the same result. The inductance L usually consists of the internal grid lead of the tube, and capacitor C may be the grid bypass capacitor. A series resonant circuit at the operating frequency is thus formed. element and the common returns of the input and output circuit, or because of excessive plate- cathode capacitance. Neutralization, where required, may be accomplished by feeding out -of -phase energy from the plate circuit to the filament circuit (figure 15A) or by inserting a reactance in series with the grid (figure 15B). For values of plate- cathode capacitance normally encountered in tubes usable in g -g service, the residual inductance in the grid -ground path provides sufficient reactance, and in some cases even series capacitance will be required. Typical tube electrode capacitances are shown in figure 16A. These can be represented by an equiva- lent star connection of three capacitors (figure 16B). If an inductance L is placed in series with C, so that a resonant circuit is formed (figure 16C), point O will be at ground potential (figure 16D) This will prevent the transfer of energy from point P to point K, since there now exists no common coupling impedance. The determination of the value of C, and L are shown in the drawing. It is apparent that when the plate- cathode capacitance of the tube is small as compared to the plate -grid and the grid- cathode capacitances, C, is a large value and the required value of inductance L is small. In practical cases the value of L is supplied by the tube . Figure 16 Tube electrode capacitances can be represented by an equivalent star connection of three capacitors. If inductance is placed in series with C, so that a resonant circuit is formed (drawing C), point 0 will be at ground potential. O CP-K CC ° CG-P t CP-K X CG-K CG-K X C-G C-K t L (2Rf)2 X Cc www.americanradiohistory.com HANDBOOK 350 Watt P.E.P. Amplifier and lead inductance, and the grid to ground impedance can be closely adjusted by proper choice of the bias bypass capacitor (figure 15B). Below a certain frequency determined by the physical geometry of the tube, neutralization may be accomplished by adding inductance to the grid return lead; above this frequency it may be necessary to series tune the circuit for minimum energy feedthrough from cathode to plate. Most tubes are sufficiently well screened so that series inductive neutralization at the lower frequencies is unnecessary, but series capacitance tuning of the grid return lead may be required to prevent oscillation at some parasitic frequency in the v.h.f. range. 29 -6 A 350 Watt P.E.P. Grounded -Grid Amplifier This section features an extremely stable, five band, grounded grid linear amplifier for sideband service. Employing the 7094 beam power tube, the amplifier provides band switched operation on all bands between 80 and 10 meters. Power output is in excess of 200 watts, and third order distortion products are better than -30 decibels below maximum two -tone signal level. High power gain, high efficiency, and low distortion can be provided economically by a high -s triode tube operating in grounded grid configuration. Beam power tubes or tetrodes (such as the 7094, 813 or 4 -250A) which can be operated as high -µ triodes make excellent grounded grid amplifiers. As a class B linear amplifier in sideband service, a triode connected 7094 with forced air cooling of the envelope can handle a conservative peak envelope -power input (p.e.p.) of 350 watts with only I750 volts on the plate and zero bias on the grids. For full input, a sideband exciter capable of an output of only 15 watts p.e.p. is required. The amplifier, complete with power supply, fits on a standard 101/2 -inch relay rack panel which may be placed within a cabinet for use directly on the operating table. Amplifier Circuit 617 The circuit of the amplifier and power supply is shown in figure 17. The plate output circuit is a bandswitching pi- network using two tapped coils and a shorting switch. The position of the taps are chosen to provide an operating Q of 15 or better on all bands with a 50 -ohm antenna load. An auxiliary loading capacitor is switched into the circuit in the 80 meter position of the bandswitch. For low impedance antennas (below 50 ohms) this capacitor should be increased in value to 1000 ppfd. The grid and screen of the 7094 tube are at r.f. ground potential. The d.c. screen return is to the cathode of the tube, and the panel meter (M1) is switched so that it is possible to read either grid current or plate current. The meter is a single- scale, 0 -300 d.c. milliammeter. A lower range meter and external shunt were not considered necessary because the normal peak grid current (80 ma.) and peak plate current (200 ma.) can easily be read on the same scale. A 1000 -ohm resistor is connected between the positive terminal of the meter and ground to prevent high voltage from appearing at the cathode of the tube in the event of switch failure. An untuned input circuit is used in the cathode for simplicity. An alternative tuned input circuit is shown. Use of the tuned circuit will result in better linearity and lower driving power requirements. If the tuned circuit is omitted, it may be necessary to "prune" the coaxial line between the exciter and the amplifier to achieve maximum driving voltage in the cathode circuit. A circuit Q of two or more is required in this tank. The power supply is a conventional full wave circuit with a choke input filter. Type 3B28 gas rectifier tubes are used in place of 866A's to eliminate the "hash" produced by the mercury vapor tubes and to permit the amplifier to be operated on its side during tests and measurements. 866A's may be used in place of the 3B28's without any circuit changes provided the amplifier is always positioned so that the tubes are vertical. The plate switch is connected in series with the filament switch so that plate power cannot be applied to the rectifier tubes until the filament circuit is energized. Filaments should be allowed to warm up for 30 seconds before plate voltage is turned on. www.americanradiohistory.com THE RADIO H.F. Power Amplifiers 618 Amplifier Because of the simplicity of the circuit it is possible to construct the amplifier and power supply on a single 12" x 17" x 3" aluminum chassis. The chassis is attached to a 101/2" relay rack panel by means of two chassis mounting brackets. The 7094 and plate tank circuit components are enclosed in a 7" x 12" x 91A " box made of 18 -gauge sheet aluminum. The front of the box is mounted flush against the rack panel, and both are drilled simulConstruction taneously for the shafts of the plate tuning and loading capacitors and the bandswitch. Half -inch wide flanges on the top and bottom of the enclosure provide good r.f. contact to the chassis and to the perforated aluminum cover plate. The small fan mounted on the rear wall of the box provides forced -air cooling for the 7094. The air intake hole is 3- inches in diameter and covered with perforated aluminum stock. Figure 17 SCHEMATIC, GROUNDED -GRID AMPLIFIER C1 -100 ;LAM., 3 kv. Johnson 100E30 (#155-10) C2-500 µpfd., 2 kv. Johnson 500E20 (#154-3) C3 -See text. 500 ppfd. mica, 1250 volts C4 section b.c. capacitor, 1100 ppld. Miller 2113 -3 -8H, 250 ma. Thordarson 20056 amp. fuse, size 3AG amp. fuse, size 3AG "slo -blo" L1 -10 and 15 meter coil: 9 turns of 3/16 inch copper tubing, 2" inside diameter, 1/2inch spacing between turns. 10 meter tap is 4/2 turns from plate end al coil. 15 meter tap is at junction between L1 and L2 L2-23 turns, B&W #3095 -1 inductor. 20 and 40 meter taps are 19 and 10 turns, respectively, from the output end of coil. Number 12 wire, 2/2" diameter L3 -18 turns # 16 wire, 1" diam., 3" long, 6 turns per inch (Air -Dux #806-T), 2.3 ph. CH1 -5 F2 -1 F1 40 meter tap (1 ph) at 9 turns, 20 meter top (0.5 ph) at 41/2 turns, I5 meter tap (0.3 ph) at 3 turns, 10 meter tap (0.15 ph) at 11/2 turns. All taps measured from ground end of coil P1,P2 -115 volt pilot lamp assembly PC turn of 1/2 -inch plate strap, 1/2-inch diameter wound about three 100 ohms, 2 watt composition resistors in parallel RFC1 -2.5 mh, 300 ma. National R -300, placed between pins 4 and 7 of tube socket RFC2 -0.225 mh, 800 ma. National R -175A RFC3 -2.5 mh, 100 ma. National R -100 Single pole, 5 position ceramic switch. Ohmite #111 or equivalent T,-6.3 volt @ 4 amp. Stancor P-4019 T2 -2.5 volt @ 10 a. Thordarson 21F02 T3-2065 -0 -2065 volts @ 200 ma. (1750 v., d.c.) Stancor PT-8315 Blowers- Cooling motor and fan. Shaded -pole induction motor, 2400 r.p.m. with 4- bladed fan, 21/2" diem. Allied Radio Co., Chicago, Ill. Part number 72P -715 -1 SI- www.americanradiohistory.com HANDBOOK 350 Watt P.E.P. Amplifier 619 Figure 18A 350 WATT P.E.P. AMPLIFIER AND POWER SUPPLY the 7094 beam tube, this compact, grounded-grid amplifier may be driven to full input with a 15 -watt The exciter. sideband complete amplifier and power supply mount behind a 101/2" relay rack panel. Panel controls are (I. to r.): meter switch, plate tuning (above) and filament switch (below), Using power bandswitch,ontennaloading (above) and plate switch (below). Figure 18B REAR VIEW OF AMPLIFIER The power supply ponents are com- grouped of the chassis. R.f. input receptacle and 115 -volt power receptacle are placed on rear apron of chassis. Antenna receptacle is mounted on rear wall of shielded enclosure. Ceramic disc capacitor is placed across meter leads directly at terminals, and leads are run in shielded braid to under- chassis area. about one end www.americanradiohistory.com 620 H.F. Power Amplifiers To hold r.f. loss to a minimum, connections between the plate tank circuit components are made of 1/4 -inch wide silver plated copper strap. Your local jeweler can probably handle the silver plating job for you. A short length of RG -8 /U coaxial line is used to make the connection between the loading capacitor and the coaxial antenna receptacle located on the rear of the enclosure. The Figure 19 TOP VIEW OF LINEAR AMPLIFIER The plate circuit is enclosed in r.f. -tight compartment bolted to the chassis deck. Power supply choke is outside rear of compartment, with plate transformer and 3B28 rectifier tubes to the right. Enclosure is covered with a piece of perforated aluminum plate for maximum ventilation. The 7094 tube is at center of enclosure, with 10 -15 meter coil between it and bandswitch. Loading capacitor is to the left, with the 20 -80 meter coil directly above it. Plate capacitor and r.f. choke are at the right. T H E R A D I O outer braid of the line is grounded at one end to the frame of the capacitor and at the other end to the shell of the coaxial receptacle. A single 8 pfd. filter capacitor is too large to fit beneath the chassis, so four 2 pfd. units are wired in parallel to provide sufficient capacity for good dynamic regulation. These capacitors, together with the filament transformers and bleeder resistors are placed in a free corner of the under -chassis area. Amplifier Tuning and Adjustment All wiring should be checked before power is applied to the amplifier. The d.c. resistance to ground of the B -plus line should be about 100,000 ohms. The amplifier is connected to the exciter and to the antenna or to a 200 watt, 50 -ohm dummy load. Mesh the plates of the loading capacitor and place the meter switch in the plate current www.americanradiohistory.com HANDBOOK 350 Watt P.E.P. Amplifier position. Turn on the plate voltage and note the resting plate current. It should be about 35 to 40 milliamperes. Apply a low level single tone signal (carrier) to the amplifier and tune the plate tank circuit to resonance. Switch the meter to indicate grid current, and advance the excitation level until the grid current reading is about 50 milliamperes. Reduce the loading capacitance, keeping the plate tank tuned, until the plate current is approximately 100 milliamperes. Increase the excitation level to obtain 80 mil:iamperes of grid current. Finally, adjust loading and tuning to obtain a resonant plate current of 200 milliamperes, keeping the grid current at 80 milliamperes. Varying the excitation level and the plate loading will permit a 2.5 -to -1 ratio between plate and grid current to be held. An exciter delivering less 621 than 15 watts may be used provided the loading is sufficiently reduced to maintain the same ratio between plate and grid current. Under voice operation, meter readings will be one -half (or slightly less) than the steady state readings indicated above. If a tuned cathode circuit is used, it is resonated for maximum grid current on each band. Figure 20 UNDER -CHASSIS VIEW OF 7094 GROUNDED -GRID AMPLIFIER Power supply components are grouped at left side of chassis. The 0.01 pfd. ceramic bypass capacitors are grouped about the socket to keep all r.f. leads short. RFC, is mounted directly on the socket between two pins. Filament transformer T, is at the right, with the rectifier filament transformer mounted to the rear wall of the chassis. Millen ceramic sockets are used for the high -voltage rectifier tubes. www.americanradiohistory.com 622 29 -7 H.F. Power Amplifiers The "Tri- Bander" Linear Amplifier for 20 -15 -10 With the advent of the trap -tuned "tri bander" beam, many amateurs are concentrating their efforts on the 20, 15 and 10 meter bands. In addition, low frequency operation is often impractical for amateurs located on small city lots and their activities must be confined to the higher frequencies, This linear amplifier is designed for the amateur whose principal interest lies in the 14 -30 Mc. spectrum. An amplifier built specially for this range can be made smaller and more inexpensively than one that covers the complete 3.5 -30 Mc. range. The unit described in this section is a one kilowatt p.e.p. class ABt cathode driven linear amplifier using two compact, ceramic 4CX300A tubes. A novel and easily built chassis- cabinet enclosure is employed, together with the inexpensive model of the Eimac air system socket. The amplifier is small enough so that it may be placed on the operating table next to the sideband exciter and receiver. Pro- T H E R A D I O visions are made for voice operation, or for operating the s.s.b. exciter without the amplifier. At 2000 volts plate potential, third order distortion products are better than -30 decibels below maximum signal input. Amplifier Circuit A high perveance tube such as the 4CX300A cannot be used in a conventional class B grounded grid circuit, as the element geometry leads to high grid current and to destructive values of grid dissipation. The distortion reduction characteristics of grounded grid circuitry, however, may be retained in an acceptable cathode driven circuit, wherein grid and screen operating potentials are applied to the tube. The schematic of this amplifier which makes use of such a circuit is illustrated in figure 22. Two 4CX300A tubes are employed, with the driving signal applied to the cathode circuit as is done in the common grounded grid configuration. Grid and screen elements are at r.f. ground, while normal Class AB' grid bias and screen potentials are applied to the tubes. Under these conditions, the power gain of the 4CX300.A is quite high; approxi- Figure 21 TRI -BANDER LINEAR AMPLIFIER FOR 10 -15 -20 METER SIDEBAND kilowatt p.e.p. linear amplifier is deThis one signed for those amateurs interested in the higher frequency DX bands. Using two 4CX300A tubes, this compact bandswitching unit is ideally suited for exciters having a p.c. p. output of about 30 watts. Panel controls are (I. to r.): Screen meter, plate meter, plate tuning, plate loading. On the left is the mode switch, Si; and on the right is the band switch, S2. Amplifier is mounted on four rubber "feet" so that cooling air may be withdrawn from under the cabinet. Geared tuning dials, switch knobs, and plate bandswitch are salvaged from surplus "TU" tuning drawers from BC191;375 transmitter. r www.americanradiohistory.com HANDBOOK Tri -Bander Amplifier E 623 AC. R.P. RY INPUT OUTPUT RFC, B+2 NV. 0 500 Hy = SIA sew*1 0= -BIAS o 500 KV T,0 1_ °RFC RFC3 u 50 250 C, 52 C2 RFC1 2 ITT RFC, 3 T 4 000 RFC. , ANT RELAYQ B+ SCR. -r -_ll _ T RFCI _ ,OOK RFC3 2K X 115 AT EACH F0rT VA, i25t) C o, RFCI SOCKET Sie 4 1 MA> E AU*.^ RFC, B CONTROL o TUNE T 1rs SWITCH RFC, 6001 CONTROL MS-1 O B CONTROL Off 2 AUX. NOTE 1 : 1. CAPACITORS -C-ARE .0011/F., 400 V, DISC CERAMIC. 2. ONE CAPACITOR "C" ON EACH GRID AND SCREEN SOCKET TERMINAL (1.1-) 3. PI LS. C T 1. 3.- 4. TUNE 5. OPERATE INDICATES PEEDTHRU CAPACITOR -11I 9 CND. Figure 22 SCHEMATIC, TRI -BANDER LINEAR AMPLIFIER -50 III,fd., 3 kv. 155 -8 (50F30), 0.075" -250 /IIId., 2 kv. Johnson spacing Johnson 155 -6 (250F20), 0.045" spacing L, -10 mete,. section: 31/2 turns, 3/16" copper tubing, wound 11/4" i.d. Adjust length to resonate with C1 25% meshed. 15 -20 meter section: S turns,, Vs" copper tubing, wound 21/4" i.d. 15 meter tap 3 turns from "cold" (output) end M, -5C d.c. milliammeter. Recalibrated to -20 to +30 ma. C1 C2 -0 M2-0 -S00 MS1 -SPST switch" d.c. milliammeter lever -type "Micro - PC-Parasitic choke. Two turns diem., wound 1/2 -inch about 47 ohm, 2 watt composition resistor RFC1 -VHF choke. Ohmite Z -144 RFC2 -44 ph., S00 ma., Ohmite «12, Z -14 RFC3 -2.5 tional mh, R -300 RY1 -DPDT, 115 tenna relay. 2C-115VA 300 ma. Na- volt coil, anAdvance AM- Si-Two pole, 5 position progressively shorting switch. Two Centralab « P -1 decks, with P -121 Index Assembly S2 Single -pole, 5 position ceramic switch from surplus "TU" tuning unit, or Centralab -- T,-6.3 volt at 6 amp. Stancor Adjust primary resistor to deliver 6.0 volts at tube sockets under load Blower-35 cubic feet per minute. 6000 r.p.m., 115 volts P -6456. a.c. Ripley «8445 -E Feedthrough capacitors -Each of the eight control leads, plus the two leads to the relay coil pass through 0.001 Aid. ceramic feedthrough capacitors. Centralab type FT -1000 Sockets: Eimac 5K -760 air socket. Place one 0.001 Aid., 600 volt ceramic capacitor from each screen terminal to ground «2550 mately 30 watts p.e.p. drive being required for full output. The amplifier plate circuit is a simple three band pi- network, designed for a circuit Q of 15. As the low frequency bands are not in- cluded, only two small self- supporting air wound coils are required. In addition, the size of the pi- network loading capacitance is considerably smaller than a capacitor necessary for all band operation. www.americanradiohistory.com 624 H.F. Power Amplifiers The amplifier is controlled by a two deck progressively- shorting switch (S1) that remotely controls the auxiliary equipment and provides the operator with a choice of "tune" or "operate" modes. All control and low voltage power leads are suitably filtered by L -C networks to suppress radiation of TVIproducing harmonics. The "Tri- bander" linear amplifier construction is novel in that no regular chassis deck is employed. The amplifier is built within an enclosure made up of two aluminum chassis, each measuring 10" x 14" x 3 ". One chassis is inverted and serves as a pan within which the components are mounted. The second chassis is placed atop the first and serves as a top shield cover. This chassis assembly is hinged along the rear edge, and opens up much in the manner of a suitcase. A single -piece front panel made of aluminum is fixed to the lower chassis. The front apron of the top section is cut away to provide clearance for the meters, switches and capacitors. When the top section is closed, the cabinet is sealed by a strip of finger stock that runs around the inside edges of the lower chassis box. A length of "piano -type" hinge fastens the rear edges of the two chassis together, and the enclosure halves are held in place by five panel bolts which screw into nut plates riveted to the lip of the lid, or top section. An aluminum partition divides the interior of the enclosure into two compartments (figure 23). The smaller compartment contains the blower motor, filament transformer, panel meters, auxiliary control relay, function switch, and power lead filters. The larger compartment contains the two 4CX300A tubes, the plate circuit pi- network components and the antenna relay. The partition is shaped to fit around the housing holding the tetrode tube sockets. As the standard air system socket with built -in screen bypass capacitor is both expensive and bulky, the smaller phenolic socket having no screen capacitor was used as an inexpensive substitute. Two of these sockets will mount atop an oscillator shield can taken from a defunct surplus "Command" transmitter. The can makes an inexpensive and r.f.tight shield for the grid and cathode components, and is mounted directly to the bottom chassis "pan." The pi- network capacitors and bandswitch are panel mounted, and the re- T H E R A D I O maining compartment area is taken up by the plate coils, r.f. choke, and the plate blocking capacitors. Antenna relay RY, is mounted within a small aluminum shield box placed at the back of the compartment. Transmitter wiring is simple and straightforward. All connections in the meter compartment are made with unshielded wire. The relay leads pass through the internal shield partition via high frequency feedthrough capacitors, and the exciter switching leads to the contacts of the relay pass through short lengths of RG -58/U coaxial line. The outer braided conductor of the line is soldered to a u.h.f. -type "hood" (Amphenol type 83 -1H) to ensure r.f.-tight connections where the cables enter and leave the amplifier compartment. The three ceramic capacitors that make up the plate blocking unit are mounted atop the plate r.f. choke, and are fastened to the main tuning capacitor by means of an aluminum strap visible in figure 23. Connection is made to the anode of each tube by means of a 1/2 -inch wide copper strap encircling the air cooler structure. Air is drawn through 1/4 -inch holes in the bottom pan by the blower, forced into the grid compartment, circulated upward through the tube socket and cooling anode, and exhausted via 1/4-inch vent holes drilled in the top lid of the enclosure. The blower motor goes on whenever the filaments of the tubes are lit. Transmitter Control Circuits and S1 controls the transmitter and auxiliary Power Supply equipment. All circuits are off in the first position. In the second position, an auxiliary circuit is completed which can turn on the station receiver or sideband exciter. The third position turns on the amplifier tube filaments and energizes the blower motor to cool the tubes. Cut -off bias is applied to the tubes to eliminate diode noise often noticed in standby operation. The fourth position applies full plate voltage and reduced screen voltage to the amplifier for tuning operations, and the fifth switch position applies full screen voltage. Cut -off bias is removed by the voice -actuated relay in the power supply. Screen and plate currents are continually monitored by the two panel meters. The screen meter is recalibrated to have an www.americanradiohistory.com Switch HANDBOOK Tri -Bander Amplifier 625 Figure 23 INTERIOR VIEW OF LINEAR AMPLIFIER The r.l. components are contained in the compartment to the right of the shield partition. Antenna relay RYA is placed in small aluminum box mounted to rear wall of cabinet directly behind antenna loading capacitor. The two 4CX300A tube sockets are mounted on top of aluminum shield can taken from oscillator coil section of surplus "command" transmitter. Micro- switch on partition removes high voltage when cover is opened. Midget relay adjacent to switch is added for auxiliary control circuits and is not required. At extreme left rear cre feedthrough capacitors mounted on aluminum plcte, with r.l. chokes beneath them. Filament transformer is in corner of compartment, in back of mode selector switch. Pi- network components are at right, with three plate blocking capacitors mounted to aluminum strip supported by plate tank capacitor. elevated zero point and reads -20 to +30 milliamperes. Under certain conditions, negative screen current can flow and it is important to monitor this sensitive indicator of amplifier operation. The power supply schematic is shown in figure 24. The high voltage supply uses 3B28 "hash-free" gas rectifier tubes and provides 2000 volts d.c. at 500 ma. and regulated 360 volts at 30 milliamperes. "Jumpers" in the base of the regulator tubes are wired in series with the primary relay circuit so that the supply cannot be energized unless the tubes are in their sockets. A smaller half -wave semiconductor supply provides operating and cutoff bias for the amplifier. The bias relay may be actuated by the voice circuit of the exciter to drop the bias to the correct amount during the time the voice circuit is energized. www.americanradiohistory.com THE RADIO H.F. Power Amplifiers 626 T, B+ 211V. CH1 RY 2A 6-1-SCR. (360 V.) (s3) 20 0.15 12 411V 311V Roo R105 GN D. (a9) VR105 n5V.'NJ SR T3 p p + 100 100 ADJ. BIAS GRID 2W 1011 RESISTOR ° 20 I 4W BIAS (a4) METER 4 2W . 7 11 01,5 V. WI (ell RY2B (a21 RY2 RY4 VOX TO PR - MANUAL JUMPERS IN VR TUBES 7 3 7 3 7 CONTACTS at S00 ma. Chicago R -65 lamp and receptacle RY2 -DPST, 11S volt coil. Potter -Brumfield MRSA, 115 volt a.c. RY3 -SPST, 115 volt coil, 20 amp. contact. Potter -Brumfield PR3AY, 115 volt, a.c. RY4 - -DPDT, 115 volt coil. Potter -Brumfield MR11A, 115 volt a.c. SR- Selenium rectifier, S00 ma. Sarkes- Tarzian h. (a7) O OPERATE CONTROL Figure 24 SCHEMATIC, LINEAR AMPLIFIER POWER SUPPLY M -500 -2.5 volts at 10 a., Chicago F -210H T2- 2900 -2300 volts each 10 kv. insulation. side of c.t. at 500 ma. 115 -230 volt primary. Chicago P -2126 v., 50 ma. Stancor PA -8421 Extra contact set of RY4 is placed in series with antenna relay control lead (17 2) and -125 RY2R CONTROL -115 volt pilot T1 T3 (a6) V.*2 X (.6) CONTROL RELAY P3 I 15 -A U TUNE' CONTROL 3 P,, ECE 0-O C11TVER SWITCH 6--ce- TO VOX CH1-6 R 1. ANT. RELAY contacts to actuate antenna relay R Y, (figure 22) by VOX circuit. The only initial adjustment is to set the operating bias level by means of the potentiometer. Initially, the arm should be set at the high potential end of the potentiometer to apply full bias to the tubes. The filaments and blower are turned on, and the high voltage and bias supply energized. Using a voltmeter, the potentiometer should be set to provide about -60 volts on the arm. The voice relay is energized dropping the cut-off Transmitter Adjustment and Tuning bias out and the potentiometer is carefully reset to provide a static plate current of 200 ma. as read on the meter. Indicated screen current (bleeder current) should be about 22 ma. When the voice relay drops out, the plate current should fall to zero. The amplifier is now fed a small exciting signal (single tone) and tuned and loaded for a maximum plate current of 500 milliamperes. Screen current should now be approximately 30 ma. (This is a total of screen and bleeder current.) The output coupling is now increased slightly so that r.f. output (as read on an r.f. ammeter, or output voltmeter) drops about 2 percent. Maximum linearity is obtained when the amplifier is slightly overcoupled. Under voice conditions, plate current peaks should reach approximately 250 ma., as read on the meter. No grid current should be read on a 0 -1 d.c. milliammeter placed across the grid current terminals in the power supply. Any flicker of grid current indicates the amplifier is being overdriven, with a consequent severe www.americanradiohistory.com HANDBOOK 813 Linear Amplifier increase in distortion. Under voice conditions, indicated screen current will be relatively constant, as actual current drawn by the screen of the tubes will be less than + or 10 ma., and this small value is swamped out by the bleeder current, which is constant at 22 ma. Low values of screen meter current (indicating that the tubes are drawing negative current) indicates excessive loading; high values of screen current indicate insufficient plate circuit - loading. Never apply excitation to this (or any other) grounded grid amplifier without all operating potentials applied to the tubes. Figure 25 THE 813 GROUNDED -GRID LINEAR AMPLIFIER Two 813's are used in this simple and effective linear amplifier. Built on a 101'2 -inch rock panel, the amplifier may be placed in a metal cabinet for desktop operation. Capable of operation on all amateur bands between 80 and 10 meters, this unit may be driven by the popular 75 to 100 watt sideband exciters. Panel controls are (I. to r.): bandswitch, plate tuning (top) and antenna loading (bottom), meter switch (top) and bias control (bottom). Front bushing of linkage shalt for switch S2 passes through panel between tuning and loading controls and is camouflaged with small knob. 29 -8 627 An 813 Grounded Grid Linear Amplifier The popular amateur s.s.b. transmitters in the 75- to 100 -watt power class provide a ready -made exciter when the time comes to add a more powerful final amplifier to the amateur station. Because tetrodes have low power drive requirements, a power dissipating device must be employed when these tubes are driven from a 100 -watt class transmitter. A suitable dissipation device is usually fragile, expensive, and difficult to construct. In addition, the tetrode tube requires bias and screen power supplies which are bulky and expensive. A grounded grid amplifier circuit provides a satisfactory solution to these problems as no power dissipating device is required, and screen and bias supplies may be eliminated. Certain tetrodes and pentodes operate well as zero -bias, grounded grid triodes, and the 813 is one of these. This tube operates efficiently in class B grounded grid service at plate poten- www.americanradiohistory.com 628 THE RADIO H.F. Power Amplifiers LI EA. 813 .001 813 57/V 7L15_2 500 L. = 1.2Kv Tool C2 KK V RFC2 II FIL. CET. 1500 0 50Ó t CAPAC ITOR RFC 50-760 RFC4 r. OUT. C1 10 KV .ó1 R 13+ 14.V. I 325 CAI ADJ. B/AS R, ALTERNATIVE TUNED CATHODE CIRCUIT .01 .01 TS2 1. 52 ITS, O Io ro 2 VOX RELAY 3 NOTES t 40M POSITIONS. OPEN /1120, POSITIONS OF BANOSW/rCN SI. CLOSED IN 80 154 10 M. 2. ALL 01 CAPACITORS ARE BOO V CERAMIC UNITS. 3. JUMPER TERMINALS / B 2 ON TERMINAL STRIP ro REMOVE BLOCK /NG B /AS. 4 115 V.1, CND Figure 26 SCHEMATIC, 813 LINEAR AMPLIFIER B- Tube -cooling motor and fan. Shaded pole induction motor, 2400 r.p.m., with 4 -blade fan, 21/2" diom. Allied Radio Co., Chicago. Part 5 72P715 C1- Two- section variable capacitor. Front section (added for 40 -80 meters): 28 -160 ppfd. Rear section: 7-50 ppfd. 0.125" spacing. Barker & Williamson. A conventional split stator capacitor may be substituted. Johnson 1;l54 -3 (100E045) is recommended. Install the switch between the stators, on the studs supporting the stator plates at the middle of the capacitor. Change length of linkage to fit new layout. C2 -1500 ppfd., 0.03" spacing. Barker & Williamson 51241. A four section, b.c. -type variable capacitor (J. W. Miller ,V.2l04) with sections in parallel may be substituted. C3 -1260 µµId. Three section b.c. -type capacitor (J. W. Miller 52113) with sections in parallel C4 -325 µµId., 0.024" spacing. Hammarlund MC -325M L, -10.5 µh. transmitting inductance. Barker & Williamson 850A. Air -Dux 5195 -2 coil may be substituted. This coil should be trimmed and topped to resonate as follows: 80 meters, 210 ppfd.; 40 meters, 105 ppfd.; 20 meters, S2 ppfd.; 15 meters, 30 µpfd.; 10 meters, 30 ppfd. Above capacities include output capacitance of tubes -10 L3 meter section: 0.44 ph. S turns 512 diam., 1" long, space -wound S turns per inch. Tapped section: 4.2 ph. 17 turns 5 16 tinned, 11/2" diom., 21/4" long, space wound 8 turns per inch. Tapped 2 (21 Mc.), 4 (14 Mc.), and 10 (7 Mc.) turns from 10 meter end of coil. B&W 53018 miniductor e., 1" -0 -1 d.c. milliammeter RFC, -0.5 mh., 300 ma. M, R -300 RFC2 -15 choke. National ampere filament choke. B&W type FC -15 RFC3 -200 ph. choke. National R -175A RFC4,3-1 mh., 300 type 800, or choke. National B&W ma. R -300 -Part of L,. An Ohmite type 111 -5, S position ceramic switch may be used with Air -Dux coil. Switch should be mounted on an insulated brocket and driven with an insulated coupling S2- Special switch. See text for details S3- Single -pole, S position ceramic. Centralab 52500 SR -130 volt, 75 ma., replacement -type selenium rectifier S, -10 volt, 10 amperes. Thordarson 21F19 volt, 50 ma. Stancor PA -8421 711,2-Insulated terminal strips. Cinch -Jones T, T2 -115 Knobs -B&W 5901 (11/2" diam., 3 req.) B&W »903 (11/16" diam., www.americanradiohistory.com 3 req.) HANDBOOK 813 Linear Amplifier 629 Figure 27 LEFT REAR VIEW OF AMPLIFIER A I á -inch thick sheet of aluminum 13 inches by 17 inches in size forms the main chassis and is fastened to the panel with chassis support brackets. Connection between plate r.f. choke, blocking capacitors, plate tuning capacitor and plate coil are made with copper strap. Plate leads from tubes to strap are made with =10 flexible braided wire. Coaxial r.f. input receptacle is next to 11S -volt line cord, and antenna receptacle is mounted on angle bracket at end of sub -chassis. Switch S, is at rear of bandswitching inductor. bals up to 3000 volts. Two 813's in parallel at 2500 volts will provide a p.e.p. input of 1500 watts (750 watts, single tone) provided cooling air is circulated about the tubes. At 3000 volts, a p.e.p. input of 2000 watts (1000 watts, single tone) may be run but the plate dissipation of the tubes exceeds the recommended maximum figure. If plenty of cooling air is used, this does not seem to shorten tube life. Under these two operating conditions, third order distortion products are better than -30 decibels below maximum power level. Amplifier Circuit The circuit of this linear amplifier is shown in figure 26. The basic amplifier employs an untuned cathode input circuit for simplicity and low cost, although an alternative tuned input configuration is shown. Improved intermodulation distortion suppression and less driving power can be gained with the use of the tuned circuit. The screen and beam -forming plates of the 813's are grounded directly at the socket. The www.americanradiohistory.com 630 THE RADIO H.F. Power Amplifiers grids are bypassed to ground and receive a small amount of negative bias from the built in bias supply. The exact bias level may be set by the potentiometer. In addition, when the connection between terminals 1 and 2 on the terminal strip is broken, the tubes are biased to cut -off to eliminate troublesome diode standby noise. When these terminals are shorted by the contacts of the voice relay, the bias is reduced to the operating value determined by the setting of the potentiometer. Separate metering of current in the grid and plate circuits is accomplished by switching a single meter (M) across shunt resistors. The 0 -1 d.c. milliammeter is converted into a low range voltmeter by the addition of the 1.2K series multiplier resistor, and the voltage drop across grid and plate shunt resistors is measured. In the grid position, the meter reads 0 -100 ma., and in the plate position it reads 0 -500 ma. A pi- network plate tank circuit is employed. Optimum plate load impedance for this circuit is about 5000 ohms, and the Q should be held to a figure of 15 or better. These requirements may be met with the specified components, or with less expensive substitutes, as outlined in the parts list. High voltage is applied to the parallel -connected 813's through the plate r.f. choke. Three blocking capacitors in parallel keep high voltage from reaching the pi- network plate tank circuit. A tapped coil and two section tuning capacitor provide nearly optimum L/C ratio on all amateur bands from 80 to 10 meters. Only one section of the tuning capacitor is in the circuit on the 10, 15 and 20 meter bands when the automatic switch S2 is open. Both capacitor sections are in parallel on 40 and 80 meters where greater maximum tuning capacitance is required, S2 being closed by a mechanical linkage from the main bandswitch, S,. A large variable pi- network output capacitor (1500 µµfd.) eliminates the need for several fixed capacitors and a tap switch to add them to the circuit as needed. The output circuit will match load impedances in the range of 50 to 75 ohms having an s.w.r. of 2/1 or less. Figure 28 RIGHT REAR VIEW OF AMPLIFIER Main tuning are mounted end -brackets !Vs -inch sheet capacitors on vertical made of aluminum. The copper nngle brackets on the plate capacitor plus U- shaped bracket on switch linkage form foreground, mounted on sub -chassis are the filament transformer, bias supply filter capacitor, high voltage terminal, and plate r.f. choke. Bottom chassis plate is drilled beneath fan to permit cooling air to be drawn into sub Sy. In chassis area. www.americanradiohistory.com HANDBOOK 813 Linear Amplifier 631 TOP VIEW REAR SUPPORT PLATE IOR CI AND C2-AY 7Y} } f ALUMINUM Y SO PANEL FRONT PLATE !BASS STRIP E- LUCITE 2 LONG +LONG LI ON U -CLIP FROM FORMED FROM it2Y=r !BASS 1 SPRING LONG FRONT SUPPORT FOR C2 CI 2 If 7 sRAS3 STRIP PANEL LONG POSITION OF LINRAGC IN 14,21 ANC 211-MC POSITIONS OF LI" Alt m;p=A=1R>=:: Vir POSITION LINKAGE AND OF 1 MC LIST O IM 3 ].]\ \ POSITIONS FRONT VIEW Figure 29 DETAIL DRAWING OF SWITCH S, LINKAGE Three ! /e" x /2" brass strips, soldered to brass shaft couplings make up the linkage arms. Plastic arm supports U -clip which closes circuit between copper angle brackets mounted on main tuning capacitor in 80 and 40 meter positions of bandswitch. Amplifier Amplifier construction is quite simple due to the utilization of standard, readily available components. The main chassis is a 14" x 17" x 1,á -inch thick sheet of aluminum fastened with its bottom surface !48-inch above the lower edge of a 101/2" x 19" aluminum relay rack panel. Only the pi-network components, meter, and meter switch are mounted to the main chassis, the remaining components being assembled on the 6" x 11" x 21/2" aluminum sub chassis. The photographs and drawings illustrate the placement of the major components. The end plates of the tuning capacitors are -inch aluminum brackets seven fastened to inches high and four inches wide (figure 30). The shaft on which the linkage for switch S2 is supported also runs between these brackets. Construction / The parts of this linkage, and assembly details are shown in figure 29. A U- shaped clip, made from spring brass or phosphor bronze, completes the connection between copper angle brackets fastened to the two stator sections on the main tuning capacitor when the bandswitch is in the 80 and 40 meter positions. The short, rotary arm on the bandswitch is adjusted so that it engages the forked arm, as shown in solid lines in the sketch when the bandswitch is in the 40 meter position. Both arms should then move up so that the forked arm is in the position indicated by the dotted lines when the bandswitch is in the 20 meter position. The rest of the plate circuit wiring is done with silver plated ', -inch copper strap. The strap is ordinary flexible copper "flashing" cut into strips and silver plated by a local utensil replating company. www.americanradiohistory.com 632 H.F. Power Amplifiers PANEL LAYOUT T H BRACKET FOR AMPLIFIER A D I O , CI 1.C2 Figure 30 PANEL LAYOUT FOR AMPLIFIER PLATE CITUNING I R SUPPORT 813 GROUNDED GRID 2 E DIA KNOBS TUNT r, HOLE DIA 22 ¡LARGER THAN METER --J-L-1 GRID ALOA ING MET -1 ri PLATE A I I( t SW S3 RI BIAS CASE I 2 LaW) L`- linkage for capacitor switch pivots on shaft located between main tuning capacitors. Drill 3/4-inch holes for this shaft, and the shafts of the capacitors, plus the meter s w itch. Aluminum is chassis -deck positioned I9 -inch above bottom edge of panel. The 2Ii3 S 4 Sub -chassis assembly and wiring is shown in figure 31. The ceramic sockets for the 813 tubes are sub -mounted on metal pillars to bring the top of the socket shell level with the under side of the top of the chassis. Under chassis wiring, with the exception of the #12 filament leads is run with #18 insulated wire. The filament choke and bias transformer are mounted on opposite walls of the chassis. A small, 115 volt blower motor and fan draws air up through 1/4-inch holes drilled in the bottom chassis plate and exhausts the air through the holes cut in the sub -chassis for the 813 tubes. Socket pins 3 and 5 are connected together and grounded to each of the two adjacent socket bolts. A jumper runs between the #4 pins, each of which are bypassed to ground by a .001 µfd. ceramic disc capacitor. Each capacitor must be a 1.2 KV type in order to carry the r.f. charging current existing in the grid circuit. In addition, a small 50 gefd, ceramic capacitor is connected between pins 1 and 4 of the tube socket nearest the filament choke. This capacitor stabilizes the amplifier in the 28 Mc. region. The 10 volt filament transformer for the 813's is placed above the chassis, as are the plate r.f. chokes and bypass capacitors. The bias filter capacitor is a can -type unit which mounts adjacent to the filament transformer. Various meter leads are brought out of the chassis via a terminal strip mounted on the side opposite the power cable and coaxial input receptacle. 4 In a TV fringe area, it may be necessary to completely shield the amplifier with perforated aluminum sheet. Amplifier harmonic content is low, and complete shielding is not necessary in an area of strong TV signals. Testing and Operating the Once construction is finished, check the filament and bias Amplifier circuits before connecting the high voltage supply to the amplifier. A power supply with provision for reducing the output to about one -half of maximum voltage is recommended, especially if the operating voltage is 2500 or higher. Connect a dummy load or antenna to the output receptacle. Caution: Never apply full excitation to this or any other grounded grid amplifier without the plate circuit tuned to resonance, and plate voltage on the stage. Damage to the amplifier tubes may result if this rule is violated. Tune-up for sideband operation consists of applying full plate voltage and (with terminals 1 and 2 on the power strip shorted) setting the bias potentiometer for 55 milliamperes of resting plate current with the meter switch set in the "plate" position. Only a few volts of bias are required, and the potentiometer arm will fall very near one end of the swing. Set the bandswitch to the frequency of the exciter and apply a small amount of driving power by injecting carrier in the s.s.b. exciter. Place the loading capacitor at full capacitance, and adjust the plate tank capacitor for resonance (minimum plate current). Apply more drive www.americanradiohistory.com HANDBOOK 813 Linear Amplifier Figure 31 TOP AND BOTTOM V EWS OF SUB- CHASSIS Filament transformer and filter capacitor are placed at left edge of chassis. 813 socket holes are frcm opposite end of chassis. Small plate choke is 2 -9, 16- inches in diameter, placed 214- inches supported on bypass capacitor terminals. Bias transformer and filament choke are mounted to underside of chassis, as is blower fan. www.americanradiohistory.com 633 634 THE RADIO H.F. Power Amplifiers to obtain about 75 ma. grid current and decrease the loading capacitor until resonant plate current rises to about 200 ma. Finally, increase the drive and increase the loading until plate current reaches 400 ma. (300 ma. at a plate potential of 3000 volts). Grid current should be approximately 100 ma. Slightly overcouple the antenna circuit until the output (as measured on an r.f. ammeter) drops about 2 percent. This will be the condition of maximum linearity. Now, switch the exciter to s.s.b. With speech, the plate current of the linear amplifier should kick up to about 135 to 150 ma.; while with a steady whistle the plate current should reach nearly 400 ma. Tune -up for c.w. operation is similar, except that the bias potentiometer is adjusted for zero (cut -off) resting plate current. With full plate voltage (2500) , the resonant plate current should be about 375 ma., with 100 ma. of grid current. At a plate potential of 3000, the plate current should be reduced to 300 ma. 29 -9 The KW -2. An Economy Grounded Grid Linear Amplifier The KW -2 sideband amplifier is designed for use with 4 -400A, 4 -250A or 4 -125A tubes, and will operate on the 80, 40, 20, 15 and 10 meter amateur bands. A pi- network output circuit is used, capable of matching 52 -ohm or 75 -ohm coaxial antenna circuits. Maximum power input is 2 kilowatts (p.e.p.) or 1 kilowatt, c.w. The amplifier may be driven by any of the popular s.s.b. exciters having 70 to 100 watts output. Full input may be achieved with the use of 4 -400A tubes, but the unit may be run at reduced power rating with 4 -250A or 4 -125A tubes. No circuit alterations are necessary when tube types are changed. The amplifier employs a passive (untuned) input circuit, and an adjustable pi- network output circuit. Air tuning capacitors are used in the network in the interest of economy and Figure 32 REAR VIEW OF AMPLIFIER PLATE CIRCUIT Sub- chassis has been removed to show ventilation holes in chassisdeck. Plate bypass capacitors are supported by t/ -inch copper strap leads. www.americanradiohistory.com HANDBOOK KW -2 Amplifier with no sacrifice in performance. The complete amplifier is housed in a TVI- suppressed perforated metal cabinet measuring 171/4" x 12" x 121/i" small enough to be placed on the operating table next to your receiver. Amplifier Circuit. The schematic of the amplifier is shown in figure 34, Two tetrode tubes are operated in parallel, cathode driven, with grid and screen elements grounded. The sideband exciting signal is applied to the filament circuit of the tubes, which is isolated from ground by an r.f. choke. The resistance of the windings of the choke must be limited to .01 ohms or less, as filament current is 30 amperes for two 4 -250A or 4 -400A tubes. Neutralization is not required because of the excellent circuit isolation afforded by the grounded elements of the tubes. The Input Circuit. The input signal is fed in a balanced manner to the filament circuit of the two tubes. Ceramic capacitors are placed between the filament pins of each tube socket, and excitation is applied to each tube through two 1250 volt, mica capacitors. The latter are employed because of the relatively high value - 635 of excitation current which may cause capacitor heating if ceramic units are employed at this point. The filament circuit is wired with #10 stranded insulated wire to hold voltage drop to a minimum. The leads from the choke to the filament transformer are run in shielded loom which is grounded to the chassis at each end of the wire. The use of shielded leads for all low voltage d.c. and a.c. power wiring does much to reduce TVI -producing harmonics. Figure 33 KW -2 LINEAR AMPLIFIER THE This two kilowatt p.e.p. amplifier uses two tubes in a grounded -grid circuit. Other tetrodes, such as the 4 -125A and 4 -250A may be used without modification to the unit. At full output, distortion products are better than -30 decibels below peak power level. Panel components are (I. to r.): Plate current meter (top) and output meter (bottom), meter switch and pilot lamp, plate tuning, band switch, and plate loading. At lower right is a tuning chart for the various bands. Chassis is bolted directly to the front panel, allowing about -inch clearance along bottom edge to permit edge of shield cage to pass between chassis and panel lip. 4 -400A / s v www.americanradiohistory.com 636 THE RADIO H.F. Power Amplifiers 4 -400A 4 -400A ANT exc N OTE: RI +METER RES /STANCE . 100.11 C2 C Figure 34 SCHEMATIC, KW -2 AMPLIFIER -0.001 pfd., 600 volt disc ceramic C,,C2,C3 -0.1 pfd., 600 -volt coaxial capacitor. Sprague "Hypass" z80P3 C4 -150 µpfd., 4500 volt. Johnson =1501345 (153 -8) Cs -SO µµId., surplus vacuum capacitor (see C The Grid Circuit. The grid circuit of this amplifier is simplicity itself. Screen terminals of both sockets are grounded to the chassis of the amplifier. The best and easiest way to accomplish this is to bend the terminal lead of the socket down so that it touches the chassis. Chassis and lead are then drilled simultaneously for a 4 -40 machine screw. Low inductance ground paths are necessary for the high order of stability required in grounded grid service. It is helpful to monitor the control grid current for tuning purposes, and also to hold the maximum current within the limits given in the data chart. Maximum grid current for the 4 -400A is 100 milliamperes. Under normal voice conditions this will approximate a peak meter reading of 50 milliamperes. Grid current can be observed by grounding the control grid of each tube through a 1 -ohm composition resistor, bypassed by a .01 pfd. disc capacitor. The voltage drop across the text) Ce -1000 text) µµId., 1250 -volt mica capacitor (see C7-1500 f.µfd. Barker & Williamson »51241 2104 pi- network coil. Air -Dux #195 2S (silver plated). Modify as follows: Strap coil: 3 turns, 13/4" diameter. Wire coil: Remove turns from free end, leaving 111/2 turns, counting from junction with tubing coil. Tap placements: 10 meters, 13/4 turns from junction of tubing coil and strop coil. IS meters, 31/4, as above. 20 meters, 11/2 turns of wire coil, counting from junction with tubing coil. 40 meters, 53/4, as above. 80 meters, complete coil in use RFC1 -30- ampere filament choke. B&W zFC- or 4 -gong b.c. capacitor. Miller L,-Kilowatt 30 RFC2- Kilowatt r.f. choke. Raypar, or B&W 2800 RFC3- v.h.f. choke. Ohmite z I -50 T1-5 volts at 30 amperes. Stancor P -6468 PC -31/2 turns z 12e, r/e" diam., 2" long. Wound around three 220 -ohm, 2 -watt composition resistors connected in parallel M1-0 -1000 ma. Triplett M2-0 -1 ma. Triplett X1- Diode, type www.americanradiohistory.com 11434 HANDBOOK KW -2 Amplifier resistor is measured by a simple voltmeter calibrated to read full scale when 100 milliamperes of grid current are flowing through the resistor. A double throw switch will permit monitoring grid current of either tube. With incorrect antenna loading, it is possible to exceed maximum grid current rating with some of the larger size s.s.b. exciters. No circuit instability is introduced by this metering technique. The Plate Circuit. Power is applied to the plate circuit via a heavy duty r.f. choke bypassed at the "cold" end by a 500 µµfd., 10 kv. "TV -type" ceramic capacitor. In addition, a v.h.f. choke and capacitor are used to suppress high frequency harmonics that might pass down the plate lead and be radiated through the power supply wiring. Two .001 cfd., 5 kv. ceramic capacitors in parallel are used for the high voltage plate blocking capacitor, and are mounted atop the plate choke. The pi- network coil is an 637 Air -Dux #195 -2S inductance, designed for service at a kilowatt level, and silver plated for minimum circuit loss. Use of the cheaper model having tinned wire is not recommended for continuous service at maximum power. The band switch is a Radio Switch Corp. #88 high voltage, ceramic switch. Figure 35 REAR VIEW OF AMPLIFIER The tube sockets are placed at the right end of the chassis, with plate r.f. choke centered between the tubes. The two plate coupling capacitors are mounted to top terminal of the choke by means of a brass strap. A "TVtype" 500 0, fd. capacitor is placed at the foot of the choke. The two panel meters are mounted orte above the other. An aluminum shield plate is placed around the rear of the meters to protect them from the strong r.f. field of the tubes. Meter terminals are bypassed, and the meter lecds are run in shielded braid. Power, control terminals, fuse and coaxial receptacles are mounted on rear apron of chassis. ) www.americanradiohistory.com 638 THE RADIO H.F. Power Amplifiers SHAFT OF SWITCH S CERAMIC PILLAR I ,TOCS Ir SHAFT OF SI SWITCH ARM to f0 -32 TOP VIEW 4 `MAMI BRASS BOLT AND NUT OF / TEFLON. PHENOLIC OTHER INSULATING TWO CONTACTS SPRING BRASS. ,ALUMINUM BRACKET, BOLTED TO OR CORNER MATERIAL OF C4 FRAME SWITCH, SHOWN IN CLOSED POSITION TOP VIEW MODIFIED INSULATED COUPLING. SWITCH ARM Figure 36 AUXILIARY PADDING SWITCH, PART OF BANDSWITCH Construction of padding capacitor switch made from parts of on insulated, flexible shaft coupler. Contacts are mode from 1/2 -inch wide strip of spring brass mounted on small ceramic insulators attached to main tuning capacitor. Contacts are shorted in 80 meter position of bandswitch. A circuit Q of 15 was chosen to permit a reasonable value of capacitance to be used at 80 meters. In this case, a 150 µµfd. variable air capacitor is employed for operation above 80 meters, and an additional 50 µµfd. parallel capacitance is switched in the circuit for 80 meter operation. The 50 µµfd. padding capacitor is the small vacuum capacitor found in the "Command" set antenna relay boxes. These capacitors seem to be plentiful and inexpensive. A satisfactory substitute would be a 50 µµfd. 5 kv. mica capacitor, also available on the surplus market. The pi- network output capacitor is a 1500 µµfd. unit. It is sufficiently large to permit operation at 80 meters into reasonable antenna loads. For operation into very low impedance antenna systems that are common on this band, the loading capacitor should be paralleled with a 1000 µµfd., 1250 volt mica capacitor. This capacitor may be connected to the unused 80 meter position of the band switch. The Metering Circuits. It is always handy to have an output meter on any linear amplifier. A simple r.f. voltmeter can be made up of a germanium diode and a 0 -1 d.c. milliammeter. The scale range is arbitrary, and may be set to any convenient value by adjusting the po- tentiometer mounted on the rear apron of the chassis. Once adjusted to provide a convenient reading at maximum output level of the amplifier, the control is left alone. Under proper operating conditions, maximum output meter reading will concur with resonant plate current dip. It is dangerous practice to place the plate current meter in the B -plus lead to the amplifier unless the meter is insulated from ground, and is placed behind a protective panel so that the operator cannot accidentally touch it. If the meter is placed in the cathode return the meter will read the cathode current which is a combination of plate, screen and grid current. This is poor practice, as the reading is confusing and does not indicate the true plate current of the stage. A better idea is to place the meter in the B -minus lead between the amplifier chassis ground and the power supply. The negative of the power supply thus has to be "ungrounded," or the meter will not read properly (figure 37) . A protective resistor is placed across the meter to ensure that the negative side of the power supply remains close to ground potential. Make sure that the negative lead between the power supply and the amplifier is connected at all times. www.americanradiohistory.com HANDBOOK KW -2 Amplifier PLATE SWITCH OR VOX RELAY 639 866'5 OR 3824.5 CONTROL RELAY T, 4 CHI 843000 V. EA. 3011 50W PRIMARY CONTROL SWITCH 100 20 W CHASSIS GROUND 15 IIS 0 ti Figure 37 SCHEMATIC, POWER SUPPLY FOR LINEAR AMPLIFIER -6 H, 500 ma. Chicago R -65 T,-3450 -2850 volts each side of center CH, 72 The Cooling System. It is necessary to provide a current of cool air about the base seals and plate seal of the 4 -250A and 4 -400A tubes. If small blowers are mounted beneath each tube socket it is possible to dispense with the special air sockets and chimneys, and use the inexpensive "garden variety" of socket. A Barber Coleman type DYAB motor and impeller is mounted in a vertical position centered on the socket, and about an inch below it. Cooling air is forced up through the socket and around the envelope of the tube. The perforated metal enclosure provides maximum ventilation, yet effectively "bottles up" the r.f. field about the amplifier. In order to permit air to be drawn into the bottom of the amplifier chassis, small rubber "feet" are placed at each corner of the amplifier cabinet, raising it about 1/2 -inch above the surface upon which it sits. The amplifier is built upon an aluminum chassis measuring 13" x 17" x 3 ". Input circuit components, power circuits, and the blower motors are mounted below the chassis, and the plate circuit components are mounted above the deck. Placement of parts is not critical, except that the leads beween the bandAmplifier Construction tap, 500 ma. 115 -230 volt primary. Chicago P -3025 -2.5 volts, 10 a. 9 kv. insulation. Chicago FH -210H switch and the plate coil must be short, heavy and direct. One -half inch, silver plated copper strap is used. The straps are bolted to the bandswitch with 4 -40 nuts and bolts. Each lead is tinned and wrapped around the proper coil turn and soldered in place with a large iron. The operation should be done quickly to prevent softening of the insulating coil material. Low resistance joints are imperative at this point of the circuit. To play safe, you can submerge the coil in a can of water, with just the top of the turns showing above the surface. This will prevent the body of the coil from overheating during the soldering process. It is also helpful to depress a turn on each side of the tap in order to provide sufficient clearance for the soldering iron. This may be done by placing the blade of a screw driver on the wire, and hitting it with a smart tap. The coil assembly is supported on four ceramic pillars, and placed immediately behind the band change switch, which is mounted on a sturdy aluminum bracket. The coil is positioned so that the taps come off on the side nearest the switch. A set of auxiliary contacts are required to switch the padding capacitor into the circuit when the bandswitch is thrown to the 80 www.americanradiohistory.com 640 THE RADIO H.F. Power Amplifiers r t urre Figure 38 UNDER -CHASSIS VIEW OF AMPLIFIER The filament transformer is mounted to the side apron, with the filament choke placed between the transformer and the tube sockets. The two blower motors are attached to an aluminum strip that holds them in position under the tube sockets, on a level with the bottom edge of the chassis. This strip is bolted to the chassis flange with flat -head bolts. The bolts holding the blowers pass through rubber grommets mounted on the strip to deaden blower noise. All low- voltage power leads run through shield braid which is grounded to the chassis by means of aluminum clamps mode from scrap material. B -plus lead is a section of RG -8 /U coaxial cable. Diode voltmeter components are mounted to a phenolic board attached to the side apron at right. meter position. A simple switch may be made up from the metal portions of an insulated coupling and a block of insulating material, such as teflon, lucite, or micarta (figure 36). The insulated disc of the coupling is removed, and an oval of insulating material is substituted. This assembly is placed on the shaft of the bandswitch. A set of spring contacts are mounted on small stand -off insulators attached to the side of the tuning capacitor and positioned so that the oval rotates between the contacts as the switch is turned. A hole is drilled in the oval, and a flat -head 8 -32 brass machine screw is passed through it. A nut is run onto the screw, and screw end and nut head are filed flat. When the www.americanradiohistory.com HANDBOOK KW -2 Figure Amplifier 641 39 PLATE TANK CIRCUIT ASSEMBLY The plate bandswitch is supported on a l's -inch thick aluminum bracket. The 80 meter padding capacitor is mounted on the front of the bracket. Silver -plated copper strap is used to make connections between the switch and the coil. Switch connections are made with 4 -40 hardware, and then soldered securely. Auxiliary padding capacitor switch may be seen on shaft of bandswitch, directly in front of bracket. Plate switch is made by Radio Switch Corp., Marlboro, N.I. 1 www.americanradiohistory.com 642 H.F. Power Amplifiers T H E R A D I O Figure 40 OPERATING CHARACTERISTICS, GROUNDED -GRID CONFIGURATION 4 -125A D.c. Plate Voltage Zero -Signal Plate Current Single -Tone Plate Current Single -Tone Screen Current Single -Tone Grid Current Single -Tone Driving Power Driving Impedance Load Impedance Plate Input Power Plate Output Power 2000 2500 10 15 3000 20 105 30 110 30 55 115 30 55 55 volts ma. ma. ma. ma. 16 16 16 watts 340 10,500 210 340 3,500 275 340 15,700 345 240 ohms ohms 1 190 145 watts watts 4 -400A (ratings apply to 4 -250A, within plate dissipation rating of 2500 3000 65 270 55 100 39 150 70 330 55 100 40 140 4500 675 435 5000 990 600 3000 4000 5000 100 120 150 700 675 540 105 170 130 104 80 150 105 106 55 115 2450 2100 3450 2700 5550 2700 1475 1870 1900 Zero -Signal Plate Current Single -Tone Plate Current Single -Tone Screen Current Single -Tone Grid Current Single -Tone Driving Power Driving Impedance Load Impedance Plate Input Power Plate Output Power 2000 60 265 55 100 38 160 3950 530 325 D.c. Plate Voltage D.c. Plate Voltage 4 -250A) volts ma. ma. mo. ma. watts ohms ohms watts watts 4 -1000A Zero -Signal Plate Current Single -Tone Plate Current Single -Tone Screen Current Single -Tone Grid Current Single -Tone Driving Power Driving Impedance Load Impedance Plate Input Power Plate Output Power switch is rotated to the 80 meter position, contact is made between the two spring arms through the body of the screw, which completes the circuit between the switch contacts. Amplifier Typical operating conditions for various tubes are tabulated in figure 40. For initial adjustment, four or five hundred volts plate potential is applied to the amplifier, and sufficient grid drive is supplied (five watts Adjustment 70 110 volts ma. ma. ma. ma. watts ohms ohms watts watts or so) to provide an indication on the plate meter. The loading capacitor is set at maximum capacitance, and the tuning capacitor is adjusted for resonance, which is indicated by the customary dip in plate current. After resonance is found full plate voltage should be applied to the amplifier, and resting plate current compared with the value shown in the table. If all is well, a carrier is applied to the amplifier for adjustment purposes. The signal may be generated by carrier injection, or by www.americanradiohistory.com HANDBOOK 4 -400A tone modulation of a sideband exciter. Caution! Do not apply full excitation to any grounded grid amplifier without plate voltage on the stage, or with the stage improperly loaded. Under improper conditions, driving power normally fed to the output circuit becomes available to heat the control grid of the tube to excessive temperature, and such action can destroy the tube in short time. Adjustable control of the excitation level is mandatory. The amplifier is now loaded to full, single tone input. (In the case of two 4- 400A's this will be 3000 volts at 333 ma., 2500 volts at 400 ma., or 2000 volts at 500 ma.) Driving power will be approximately 30 watts per tube. Under these conditions, power input will be 1000 watts p.e.p. for sideband operation. To properly load the amplifier for 2 kw. p.e.p, operation it is necessary to have a special test signal. Tuning of this (or any other linear amplifier) is greatly facilitated by the use of an oscilloscope and envelope detectors. Even with two -tone or carrier input signal, however, it is difficult to establish the proper ratio of grid drive to output loading. In general, antenna coupling should be quite heavy: to the point where the power output of the amplifier has dropped about two percent. This point may be found by experiment for power levels up to 1 kw. p.e.p. However, since neither this amplifier, nor most power supplies, are designed for continuous carrier service at two kilowatts and since this average power level is illegal, some means must be devised to tune and adjust a "legal" two kilowatt p.e.p. linear amplifier without exceeding the limitations of the amplifier, and without breaking the law. A proper test signal having high peak to average power ratio will do the job, permitting the amplifier to run at less than a kilowatt d.c. input while allowing the 2 kw. peak power level to be reached. This type of signal can be developed by an audio pulser, such as was described in QST magazine, August, 1947 ( figure 41) The duty cycle of this simple pulser is about 0.44. This means that when the amplifier is tuned up for a d.c. indicating meter reading 800 watts, using the pulser and single tone audio injection, the peak envelope power will just reach the 2 kw. level. An oscilloscope and . Amplifier 643 6J5 0010 AUDIO INPUT PULSED AUDIO OUTPUT MEA /r.P-3045 OR FOUI VALENT Figure 41 AUDIO PULSER FOR HIGH POWER TUNE -UP OF AMPLIFIER This simple audio pulser modifies the audio signal to the sideband exciter so that it has a high peak -to- average power ratio. Amplifier may be thus tuned for two kilowatt p.e.p. input without violating the one kilowatt maximum steady state condition. audio oscillator are necessary for this test, but these are required items in any well equipped sideband station. Loading and drive adjustments for optimum linearity consistent with maximum power output may be con ducted by this method. 29 -10 A Pi- Network Amplifier for C -W, A -M, or SSB This all- purpose amplifier covers the 3.529.7 Mc. range, and is designed for one kilowatt c.w. or s.s.b. operation, and 825 watts input plate modulated a.m. service. Using a single 4 -400A tetrode tube, this grid- driven amplifier may be driven by an exciter having a power output of approximately 15 watts. Two mechanical designs are discussed, one using variable vacuum tuning capacitors, and the other employing the less expensive variable air capacitors. The latter design is highly recommended as an inexpensive and foolproof amplifier for the amateur wishing to go high power on a lean purse! Amplifier Circuit The schematic of the amplifier is shown in figure 43. Bandswitching is employed in the grid and plate circuits, and the tetrode tube is www.americanradiohistory.com 644 H.F. THE RAD Power Amplifiers Figure 42 4 -400A ALL -BAND AMPLIFIER This compact amplifier is designed for operation in the 3.5 -29.7 Mc. ronge. Using bandswitching in the grid and plate circuits, the unit is capable of a full kilowatt input on c.w. and s.s.b., and 825 watts a.m. phone. The amplifier employs variable vacuum tuning capacitors, but an alternative design uses inexpensive air capacitors. Panel controls are (I to r.): plate current meter (top), grid bandswitch (center), and grid tuning (bottom). Screen current meter (top), plate tuning (center), and plate loading (bottom). Grid current meter (top), plate bandswitch (center), and filament switch and pilot lamp (bottom). www.americanradiohistory.com ;O HANDBOOK 4 -400A Amplifier 645 EA 300 ,0 KV 4- 250A /4 -400A RF L 2 OUT. RFC ISO CS 2 470 S1A Ext. 1500 S4 L1 27.1V. 60 SOO SEE RFC, NOTE 110MV NOTE. C IS .01 Ur CORAM IC, 600 VOLT. Hn B+HV SCLr MODULATION CIRCUIT OC FOR SCRCCN LOAD. X OMONE 12 ,011 zwll --IHII 470 C -BIAS B MV 115V."1. Uri O +SCR CND. +BIAS -SCR. Figure 43 SCHEMATIC, 4 -400A AMPLIFIER C1 -140 ppfd. Hammarlund C2- Neutralizing capacitor. APC -1408 10 µpfd. Millen #15011, or Johnson N -250 UCS -250 variable Jennings C3 -250 µµid. vacuum capacitor. Johnson 250070(153 -13) C4 -1500 ppfd. Jennings UCSL -1200 variable vacuum capacitor. J. W. Miller #2I04 air capacitor may be substituted L1 -50 turns, #24, 13/4" long, 3/4" diam. Tap S, 8, 13, and 25 turns from grid end. Wound on ceramic form. Link coil is 4 turns #18 insulated wire, wound on "cold" end of LI, tapped at center of winding L2- Barker d Williamson #850 pi- network inductor. 80 meters, 13.5 ph.; 40 meters, 6.S ph.; 20 meters, 1.75 ph.; 15 meters, 1.0 ph.; 10 meters, 0.8 ph. -0 -50 d.c. milHammeter M2 -0 -100 d.c. milliammeter M3 -0 -800 d.c. milliammeter PC-4 turns, 1" diam. wound about MI four 220 ohm, 2 watt composition resistors in parallel RFC,,s -2.5 mh. National R -100 RFC2 -BSW #800 plate choke, or National R -175A S1 pole, S position ceramic switch. Centralab 2002 volts @ 15 amperes. Triad F -9U T1 Blower-Shaded pole induction motor, 2400 r.p.m. 4 blade fan, 21/2" diam. Allied Rodio Co., Chicago, part number 72P -715 Counter dials: Grath Mfg. Co. -2 -5 neutralized to achieve maximum stability of operation. Link coupling from the external exciter is used, and a tuned grid circuit offers maximum rejection to any spurious harmonics or unwanted emissions of the exciter. Capacitive bridge neutralization is employed, with a 250 ¡yid. mica capacitor forming the ground leg of the bridge in the grid circuit. Each screen terminal of the tube socket is bypassed to ground with a low inductance high voltage ceramic capacitor, and the screen power lead is harmonic filtered by a simple R -C network. Grid and screen currents are separately metered. To aid circuit stability in the region of v.h.f. parasites, one leg of the filament is grounded, and the opposite terminal is bypassed to ground at the tube socket. In addition, simple parasitic chokes are used in the grid and plate circuits as a safety measure. The plate circuit is the popular pinetwork configuration, and will match 50or 75 -ohm antenna loads having an s.w.r. of less than 2 to 1. Amplifier plate current is metered in the B -minus lead to the power supply in order to remove the meter from the high potential B -plus circuit. By returning the bias and screen supplies to the cathode circuit (ground) the plate meter reads only the true plate current and not the cathode current, which is the sum of grid, screen, and plate currents. The reader is referred to the discussion of this subject in a previous section of this chapter. www.americanradiohistory.com THE RADIO H.F. Power Amplifiers 646 Figure 44 TOP VIEW OF 4 -400A AMPLIFIER R.f. circuits atop the chassis are enclosed in ventilated box made of perforated aluminum. Band switching inductor is at the right, with coaxial antenna receptacle directly to the rear, mounted on aluminum plate. To left of variable vacuum capacitor is the disc -type neutralizing capacitor. Plate r.f. choke is directly behind tube. Panel meters are isolated from r.f. field by aluminum sub -panel. Amplifier The amplifier is constructed upon an aluminum chassis measuring 15" x 17" x 4 1/2". Standard, TVI -proof construction is used, as outlined in the Workshop Practice chapter of this Handbook. The above -chassis circuitry is enclosed in a perforated aluminum Construction enclosure measuring 13 t/4" x 17" x 9 ". The frame of the enclosure is made of t/2 -inch aluminum angle stock, with corner gusset plates. Perforated sheets form the sides and top and are held in position with sheet metal screws spaced about three inches apart along the edges of the material. A sub -panel made of I/8-inch aluminum is placed about 13/4 www.americanradiohistory.com HANDBOOK 4 -400A inches behind the main panel. The area between the two panels is taken up by the three meters, and the gear drive system for the grid bandswitch. The panels are held in position by metal spacers located at the extreme top corners of the assembly. Placement of the major components may be seen in the photographs. The pi- network tuning capacitors are centered on the panel, with the bandswitch controls placed symmetrically about the tuning capacitor. Below deck the output loading capacitor is contained within a small shielded compartment formed from sheet aluminum. As the grid input circuit is adjacent to this capacitor, it is important that :here be no leakage of r.f. energy from input to output circuits. The bottom plate of the chassis is a solid piece of aluminum, with a 4 -inch hole cut in it directly below the blower for the tube socket. The hole is covered with perforated aluminum stock, and the bottom plate is firmly bolted to the chassis lip, and also to the flanges of the box screening the output loading capacitor. An "r.f.- tight" box thus surrounds the capacitor. Connection between the capacitor and the pinetwork circuit above the deck is made via a ceramic feedthrough insulator mounted in the deck. The blower motor is mounted in a vertical position below the ceramic tube socket (figure 44A ). A strip of aluminum supports the motor between the lip of the chassis and a lip of the capacitor compartment. The bracket is mounted with flat-head bolts, and the motor bolts are run through rubber grommets mounted in the strip. The power leads to the motor, as well as all other low voltage power wiring beneath the chassis, are run in shielded braid with the lead bypassed to the braid at each end of the run. The grid circuit components are mounted to an aluminum plate spaced away from the panel by four aluminum posts. The grid capacitor is driven by two flexible couplings from the tuning dial, which is positioned on the panel below the bandswitch and meter. The grid bandswitch is driven from atop the chassis by means of two right -angle gear drives. One drive is below the chassis and the second is placed in the meter compartment behind the bandswitch dial. Amplifier 647 Placement of the major plate circuit components may be seen in figure 44. The tuning capacitor is centered on the chassis with the tube and neutralizing capacitor on one side, and the plate tank inductor on the opposite side. The ceramic plate circuit coupling capacitors are mounted between two aluminum plates, forming a "sandwich" supported on one side by a 1/2-inch wide copper strap from the plate r.f. choke, and on the other side by a similar strap affixed to the plate tank capacitor. The bias and screen supply described in the next section of this chapter may be used for all- purpose amplifier operation. Screen protective relay RY1 should be adjusted to cut out at a maximum screen current of 50 milliamperes. If sideband operation is not contemplated, it is possible to eliminate the voltage regulator tubes in the screen supply and substitute a simpler unit that will provide 400 volts d.c. at 50 milliamperes. This will be suitable for either phone or c.w. operation. For the former, it is necessary to allow the screen to "self- modulate" itself to obtain 100 percent plate modulation. This is done by inserting a 10 -henry filter 100 ma. choke in the screen lead at the point marked "X" (figure 43) . The choke is shorted out for c.w. operation. Bias and Screen Supply Use of Air In order to reduce the cost of the amplifier, it is possible to substitute air capacitors for the variable vacuum units. A Johnson #250D70 (153 -13) will serve as the plate capacitor, and a four gang b.c. -type capacitor, such as the J. W. Miller #2104 will replace the vacuum output capacitor. In addition, the inexpensive Air -Dux inductor and the ceramic switch described in the "KW -2" amplifier may be used as a substitute for the more expensive bandswitch assembly shown here. Capacitors Amplifier Tuning and Adjustment The amplifier should be neutralized in the manner described in the next section of this chapter. Proper neutralization is indicated during operation of the amplifier by detuning the plate tuning capacitor a small amount each side of resonance. The point of www.americanradiohistory.com 648 H.F. Power Ampli=fiers THE RADIO Figure 44A LAYOUT OF UNDER -CHASSIS COMPONENTS The pi- network loading capacitor is mounted on angle plates within the shielded compartment at center. The grid circuit components a-e at the left, in fient of blower fa- and motor. The filament transformer is mounted to the wall at right side of chassis. Shielded wire .s used for all low- voltage power leads. www.americanradiohistory.com HANDBOOK Kilowatt Amplifier minimum plate current should coincide with the point of maximum grid current. If grid current increases when the plate circuit is tuned either side of resonance, the setting of the neutralizing capacitor should be varied slightly until the two readings coincide at one capacitor setting. The bias supply is adjusted to provide approximately -120 volts of cut -off bias. Full screen voltage may be applied as long as cutoff bias is on the stage. Full excitation, however, should never be applied in the presence of screen voltage unless full plate voltage is on, and the amplifier is properly loaded. Screen current is a very sensitive indicator of proper operation. High values of screen current point to insufficient antenna loading, or to excess drive. Low screen current indicates excessive antenna loading or insufficient drive. If the plate current seems normal, the drive level should be adjusted to provide proper screen current. 29 -11 Kilowatt Amplifier for Linear or Class C A pair of 4 -250A or 4 -400A tetrode tubes may be employed in a pi-coupled amplifier capable of running one kilowatt input, c -w or plate modulated phone, or two kilowatts p.e.p. for sideband operation. Correct choice of bias, screen, and exciting voltages will permit the amplifier to function in either class A, B, or class C mode. The amplifier is designed to operate at plate potentials up to 4000 volts, and excitation requirements for class C operation are less than 25 watts. A bandswitching type of pi- network is employed in the plate circuit of such an amplifier, shown in figure 45 The pi- network is an effective means of obtaining an impedance match between a source of r.f. energy and a low value of load impedance. A properly designed pi- network is capable of transformation ratios greater than 10 to 1, and will provide approximately 30 decibels or more attenuation to the second harmonic output of the amplifier as compared to the desired signal outpiat. Since the second harmonic level of the amplifier tube may already be down some 20 db, the actual second harmonic output of the network will be down perhaps 50 db from the fundamental power level of the transmitter. Attenuation of the third and higher order harmonics will be even greater. Operation Figure 45 GENERAL PURPOSE AMPLIFIER OPERATES IN CLASS A, 8, OR C MODE This kilowatt employs a amplifier pair 649 of 4- 250A's in a pi- network circuit. Mode of opera- tion may be set by selection of proper screen and bias voltages. Grid, plate, and screen current meters are mounted on plastic plate behind panel cut -out, and tubes are visible through shielded panel opening. Across bottom of panel (left to right' ore bandswitch, grid tuning, plate tuning, loading, and primary power control circuits. Plate tuning knob is attached to small counter dial. www.americanradiohistory.com 650 H.F. Power Amplifiers Le 4 -250A 4 -250A 4 -4004 4 -400A L2 L3 L EACH 001 5NV S S OUTPUT 1 J2 H( L4 Ls RFCI RFC 2 5ó1 1 500 20NV NOTE: SCREEN BYPASS CAPACITORS ARE CENTRALAB TYPE 838 INPUT JI 01 /M3 y 01 +T Hl1 L TS1 -BIAS CON- rROL 115V +SCREEN GN0 ti 115v. B+2500- ti 3500 Figure 46 SCHEMATIC, GENERAL PURPOSE KILOWATT AMPLIFIER S -Two pole, 6 position ohm, 2 watt PC-47 T -S volt, 20 ampere. -100 µp /d. Hammarlund HF -100 Cr-200 µµId., 10KV variable vacuum capacitor. Jennings UCS -200 Cs-1500 µtd., variable Cardwell capacitor. CI 8013 C.-Neutralizing capacitor, disc. Millen 15011 C1 -300 µµtd., 1250 volt L1 -L,, -See coil mica, table composition wound with 6 switch. Two Centralab PA -17 decks, with PA301 index assembly resistor turns = 18e. RFC: -2.5 mh. choke. R -100 National -Two S: RFC.-Heavy duty, wide band r.i. choke. Barker 8 Williamson type 800 RFC -VHF ite 51 choke. Ohm- Z -144 The peak voltages encountered across the input capacitor of the pi- network are the same as would be encountered across the plate tuning capacitor of a single -ended tank used in the same circuit configuration. The peak voltage to be expected across the output capacitor of the network will be less than the voltage across the input capacitor by the square root of the ratio of impedance transformation of the network. Thus if the network is transforming from 5000 ohms to 50 ohms, the ratio of impedance transformation is 100 and the square root of the ratio is 10, so that the voltage across the output .capacitor is 1 /10 that across the input capacitor. A considerably greater value of maximum capacitance is required of the output capacitor than of the input capacitor of a pi- network when transformation to a low impedance load is desired. For 3.5 Mc. operation, maximum values of output capacitance may run from Stancor M: pole, 6 position high voltage switch. Communication Products Co. type 88 two gang switch -Four pole, three position switch. Centra lab P -6492 -0 50 Triplett - - 150 Triplett ma. ma. d.c. d.c. -0 750 ma. d.c. Triplett Gears-2 required. Boston Gear CG -465 and M - =G-466 500 µµfd. to 1500 µµfd., depending upon the ratio of transformation. Design information covering pi- network circuits is given in an earlier chapter of this Handbook. Illustrated in this section is an up -todate version of an all -band pi- network amplifier, suited for sideband or class -C operation. The unit is designed for TVI -free operation over this range. Circuit The schematic of the general purpose amplifier is shown in figure 46. The symmetrical panel arrangement of the amplifier is shown in the front view (figure 45) and the rear view (figure 47) . A 200 µµfd. variable vacuum capacitor is employed in the input side of the pi- network, and a 1500 µµfd. variable air capacitor is used in the low impedance output side. The coils of the network are switched in and out of the circuit by a two pole, five Description www.americanradiohistory.com Figure 47 REAR VIEW OF GENERAL PURPOSE AMPLIFIER WITH SHIELD REMOVED The pi- network circuit is built from an inex- pensive high voltage rotary switch, and five inductors. The switch is panel driven by a gear and shaft system shown in figure 38. Variable vacuum capacitor is mounted vertically between the tubes, directly in back of the plate r.f. choke. Neutralizing capacitor is at right, connected to plates of tubes with a wide, silver plated copper strap. Meters are enclosed by aluminum shield partition running the width of the enclosure, with conduit carrying meter loads to under- chassis area at left, front of chassis. Metal shells of tube bases are grounded by spring contacts. position high voltage ceramic rotary switch. Each coil is adjusted for optimum circuit Q, resultine in no tank circuit compromise in efficiency at the higher frequencies. A close -up of the tank circuit is shown in figure 47. The plate blocking capacitor is made of two .001 µfd., 5 kv. ceramic capacitors connected in series. Special precautions are taken to insure operating stability over the complete range of amplifier operation. The screen terminals of each tube socket are jumpered together with Ye" copper strap and a parasitic choke (PC) is inserted between the center of the strap and the screen bypass capacitor. In addition, sup- O u pressor resistors are placed in the screen leads after the bypass capacitor to isolate the sensitive screen circuit from the external power leads. A third parasitic choke is placed between the grid terminals of the tubes and the tuned grid circuit. The five coils of the grid circuit are enclosed in a small aluminum shield placed adjacent to the tube sockets (figure 48 and figure 50). The amplifier is neutralized by a capacitive bridge system consisting of neutralizing Figure 48 PLACEMENT OF PARTS IN UNDER CHASSIS AREA Grid tuned circuit is en- closed in separate enclosure at left. Bandswitch projects out the rear of case, and is gear driven by same shaft that actuates the plate band switch. Switches are driven through right -angle gear drives and gears. Output capacitor of pinetwork is shielded from under -chassis of rest components. The screen terminals of each tube socket are strapped together with ribbon, and 3/e" copper inductance screen low is capacitor bypass socket to grounded mounting bolt. Screen parasitic choke mounts between strap and ca- pacitor terminal. All power leads beneath the chassis are run in shield- ed braid, grounded to at convenient chassis points. B -plus lead is made of section of RG8,'U coaxial cable, with outer sheath and braid removed. www.americanradiohistory.com H.F. Power Amplifiers 652 COIL TABLE T H FIGURE 49 KILOWATT AMPLIFIER FOR GRID COILS Li- (SO METERS) : L2 -(40 METERS I: L3-(20 METERS): N24 E, 3/4' O /A., 1" LONG ON AMPNENOL POLYSTYRENE FORM. 40 TURNS 24 3O TURNS E, .7/4"DIA., 3/4- LONG ON AMPNENOL POLYSTYRENE FORM. 72 TURNS, 82W 3071 MINIDUCTOR, L4 -(1S METERS): 7TURNS, 88 3O1O MINIDUCTOR, S/4" OM., 3/4' LONG. 3/4' DIA., 7/e' LONG. LE -(10 METERS): 3 TURNS. AS ABOVE. ALL COILS HAVE 3 TURN LINKS MADE OF HOOKUP WIRE. PLATE COILS LS- METERS) 77 TURNS !O, 3 L7- (40 METERS): IO TURNS IO, L! -lao METERS): P (SO : -O O., 4 TURNS PER INCH -AIR -DUX LP -(1S METERS): L10 -00 METERS): TURNS, 2 1/2' 3/7" O.O., 3" OD., COPPER TURING, LONG. 1/4" COPPER TURING. /4- COPPER TURING. 7 TURNS, 1/4" 0.0., .1" LONG. S TURNS, I I/4" 0.0., - TURNS PER /NCH 3" 2 2 5 3 "LONG. capacitor G and Cr., the grid circuit bypass capacitor. Screen voltage may be removed for tune -up purposes by control switch 5., section B. The screen circuit is grounded in the "off" and "fil" positions by means of switch section C. Amplifier The complete amplifier is built upon an aluminum chassis measuring 13" x 17" x 3" and has a 14" standard relay rack panel. The Construction Figure 50 GRID TANK CIRCUIT ASSEMBLY Coils are mounted to the ceramic switch decks by their leads. A small aluminum plate attached to rear of the switch assembly rods supports grid tuning capacitor which projects out rear of shielded enclosure. Entire assembly may be pre -wired before placing in enclosure. E R A D I O grid circuit components are mounted within an aluminum box measuring 3" x 4" x 4 ". Plate loading capacitor G, r.f. choke RFC -1, and output connector J2 are placed within an enclosure measuring 6" x 6" x 3", made up of aluminum angle sections and sheet material. The plate circuit shielding is made of Reynold's "Do- it- yourself" aluminum stock, available at most hardware stores. Layout of the major components can be seen in figure 47. The two tube sockets are placed directly behind the panel opening, with the plate r.f. choke between them, and the variable vacuum capacitor is mounted vertically to the chassis directly behind the sockets, on the center line of the chassis. To the right of the sockets is neutralizing capacitor C.. The high voltage ceramic coil switch SA-B is placed directly behind the vacuum capacitor, mounted in a vertical position. The variable vacuum capacitor is panel driven by a counter -type dial, through a miniature right angle gear drive, as seen in the under chassis view (figure 48). The plate and grid band switches are ganged and switched in unison by means of a shaft acting through two right angle gear drives and two bevel gears. Both circuits are thus switched by the "Band switch" control located in the lower left corner of the front panel. It is necessary to apply forced air to the sockets of the amplifier tubes. A large 115 volt a.c. operated blower is therefore mounted in the center of the bottom shield plate. The under -chassis area is thus pressurized and the majority of the air escapes through the socket ventilation holes located near the pins of the tubes. All wiring beneath the chassis (with the exception of the filament leads) is done with 5KV insulated wire, encased in metallic braid which is grounded to the chassis every inch or so. The B -plus wiring from the high voltage terminal to the plate current meter is done with a section of RG -8 /U coaxial line from which the outer braid has been removed. A similar piece of line is run from the ,peter to the plate r.f. choke, RFC -2. The three meters are mounted upon a lucite sheet placed behind a second lucite sheet mounted behind a cut-out in the front panel. The meters are shielded from the plate circuit of the amplifier by an aluminum enclosure that covers the wiring and meters, running the full length of the chassis. The meter leads pass through the plate circuit area via a short length of 1,1,-inch aluminum conduit that is threaded www.americanradiohistory.com HANDBOOK Kilowatt Amplifier 4-250A /4-400A OPERATING CHARACTERISTICS (2 TUBES ) MODE ITEM PLATE VOLTAGE 55861 33682 PHONE C.W. 3000 3000 2500 3000 400 330 600 S00 400 500 24 1.0 60 70 GRID BIAS -110 -60 PROTECTIVE BIAS -110 -60 -200 -120 -200 -120 PLATE CURRENT (MA.) 110 SCREEN VOLTAGE SCREEN CURRENT (MA.) GRID CURRENT (MA.) POWER OUTPUT(WATTS) Ti -420 260 -440 0 0 20 20 600 700 770 600 5R4 -GY CH, R1 5K sow RY1 111 2 o 1 loF nv 5 T U VR -1150 5GR 5 VR -150 1.3 v. 1oó VR 1s OR V 115V. R- 150 VR T2 /W ID -90 9 GND 5 OR VR -150 106.3V. `\\\ V. ( (( 5Y3-GT * 20LIF I 450 5K/25w R3 + R4 R2 6 I LW e St -BIAS Figure 51 OPERATING DATA AND SCHEMATIC, SCREEN AND BIAS SUPPLY volts of 150 ma. and -0- 410 -870 T:- 870 -410 volts, ma. 60 5 Stancor P Te- 235 -0 -235 CH CH 2 o., 6.3 v. 3.5 a. -8307 volts at 40 ma. Stanco, PC- 8401 -7 henry -7 henry at 150 ma. Stancor C -1710 at 50 ma. Stancor C -1707 relay, adjustable 100 -250 ma. insulated from chassis. RY,- Overload Note: J, is at each end and bolted to the chassis and the meter shield. Plate circuit wiring above the chassis is done with 1/2-inch silver plated copper strap. After the amplifier is wired and checked, it should be neutralized. This operation can be accomplished with no power leads attached to the unit. The tubes are placed in their sockets, and about 10 watts of 30 Mc. r.f. energy is fed into the plate circuit of the amplifier, via the coaxial output plug J,. The plate and grid circuits are resonated to the Amplifier Neutralization 653 frequency of the exciting voltage with the aid of a grid -dip meter. Next, a sensitive r.f. voltmeter, such as a 0 - 1 d -c milliammeter in series with a 1N34 crystal diode is connected to the grid input receptacle (J1) of the amplifier. The reading of this meter will indicate the degree of unbalance of the neutralizing circuit. Start with a minimum of applied r.f. excitation to avoid damaging the meter or the diode. Resonate the plate and grid circuits for maximum meter reading, then vary the setting of neutralizing capacitor G until the reading of the meter is a minimum. Each change in G should be accompanied by re- resonating the grid and plate tank circuits. When a point of minimum indication is found, the capacitor should be locked by means of the auxiliary set screw. Complete neutralization is a function of the efficiency of the screen bypass system, and substitution of other capacitors for those noted in the parts list is not recommended. Mica, disc -type, or other form of bypass capacitor should not be substituted for the units specified, as the latter units have the lowest value of internal inductance of the many types tested in this circuit. The amplifier requires -60 to -110 volts of grid bias, and plus 300 to 600 volts of screen potential for optimum characteristics when working as a class AB1 linear amplifier. Screen voltage for class C operation (phone) is 400 volts. The voltage may be raised to 500 volts for c.w. operation, if desired, although the higher screen voltage does little to enhance operation. Approximately -120 volts cut -off bias is required for either phone or c -w operation. A suitable bias and screen power supply for all modes of operation is shown in figure 51, together with an operating chart for all operating voltages. The supply furnishes slightly higher than normal screen voltage which is dropped to the correct value by an adjustable series resistor, R1. This series resistor is adjusted for 30 milliamperes of current as measured in meter jack J1 when the supply is disconnected from the amplifier. Series bias resistor R2 is adjusted for the same current in jack J. under the same conditions. The value of protective bias may now be set by adjusting potentiometer R3. Additional bias is required for class C operation which is developed across series resistor R.. Switch S1 is open for class C operation and closed for sideband operation. It is imperative that the screens of the tetBias and Screen Supply www.americanradiohistory.com 654 H.F. Power Amplifiers rode amplifier tubes be protected from excessive current that could occur during tuning adjustments, or during improper operation of the amplifier. The safest way to accomplish this is to include an overload relay that will open the screen circuit whenever the maximum screen dissipation point is reached. Two 4250A tubes or 4 -400A tubes have a total screen dissipation rating of 70 watts, therefore relay RY-1 should be adjusted to open the screen circuit whenever the screen current reaches approximately 100 milliamperes. 29 -12 A 2- Kilowatt All -Band P.E.P. Amplifier T H E R A D I watts p.e.p. output in sideband service. Maximum grid dissipation of the 4CX1000A is zero watts. The design features which make the tube capable of maximum power operation without driving the grid into the positive region also makes it necessary to avoid positive grid operation. This efficient amplifier covers the 3.5 -29.7 Mc. amateur range and may be driven by any modern sideband exciter having a power output of 75 watts, p.e.p. In addition to sideband operation, the amplifier may be used as an a.m. linear, providing a carrier power of about 350 watts. Circuit Description Described in this section is a deluxe all band linear amplifier suited for s.s.b. or c.w. operation up to the maximum legal power limit. A 4CX1000A ceramic power tetrode is employed in a basic passive grid circuit shown earlier in this chapter in figure 11C. The 4CX1000A is a ceramic and metal, forced air -cooled, radial beam tetrode with a rated maximum plate dissipation of 1000 watts. It is a medium voltage, high current tube specifically designed for Class AB1 r.f. linear amplifier service where its high gain and low distortion characteristics may be used to advantage. At the maximum rated plate voltage of 3000, the tube is capable of 1680 O The circuit of this all band amplifier is shown in figure 53. A resistance loaded, passive grid configuration is employed in conjunction with a pi- network output circuit. Grid drive requirements are about 60 volts peak, developed across resistor R1 which has a value of 50 ohms. This corresponds to approximately 72 watts p.e.p., all of which is dissipated in the grid resistor. Average power dissipated in this resistor is about 30 watts under voice waveform conditions. It is possible to tune up and adjust the exciter with the plate and screen voltages removed from the 4CX1000A, using this resistor as a dummy load. The amplifier plate circuit is the popular pi- network configuration employing a tapped Figure 52 DELUXE 4CX1000A SIDEBAND AMPLIFIER Constructed in a desktop cabinet, this 5 -bond sideband amplifier is rated at 2 kilowatt p.e.p. level. Panel controls are (I. to r.): meter switch, plate tuning (top), filament and plate switches and pilots (center), plate loading control (bottom) and bandswitch. www.americanradiohistory.com HANDBOOK 4CX 1000A Amplifier EA 4CX1000A 655 500 RF OUT RF .001 zKV IN 1 R FC 2 = 0.1 tKx42 500. SEE TEXT W E' E R ADJ. B 5000 /AS 2w 15+ HV R F OUTPUT E' rDW ,0K MULTIMETER SWITCH S1 S,* ,ow A - 0 -, MA., GRID BSIB C l 0 -1 - 0 MA., PF OUT. -40 MA , SCREEN D- 0- 500V., 111, SCREEN e-0 -5KV., DA PLATE 0 -1 NOTE 2 -BIAS A.C. PIL B+ HEUT SW ON 5V. N B B+ -500 µµtd., 5 kv. Jennings Radio Co., type UCSL -2000 NF,td., 2 kv. Jennings Radio Co., type UCSL 8 Williamson x.852 turret RFC,-2.5 mh. National R -100 RFC2- Transmitting type r.f. choke. National R -175A T, -6.0 volts @ 11 a. Stancor P -6463 Blower -S0 cu. ft. min. Ripley »81 or equivalent C2 80P GND. SCR. Figure 53 SCHEMATIC, 4CX1000A AMPLIFIER C1 B- 0.1 CAPACITORS ARE SPRAGUE -3 NYPASS L,- Barker coil and variable vacuum capacitors. A simple diode voltmeter is used to monitor the r.f. output voltage of the amplifier. The network is capable of matching antenna loads of 50 -75 ohms, which exhibits an s.w.r. of less than 2/1. Metering and Control Circuits. The amplifier unit contains two panel meters (figure 54) . A 0 -1 d.c. ammeter placed in the B -minus leg of the power supply serves as a plate meter. The second meter is a 0 -1 d.c, milliameter connected as a multi -purpose indicator. Panel switch S1 places the meter across shunt and multiplier resistors in various circuits. The basic control circuits are shown in figure 55. A "fail- safe" design utilizes control relays energized from low voltage d.c. supplies. If one of the supplies fails, or a relay becomes inoperative for any reason, the 4CX1000A tube is protected from abuse. Inexpensive 115 volt a.c. relays are used, which operate satisfactorily from a d.c. source of about 30 volts. Series resistors may be used with the relay coils to provide the correct pull -in voltage. Relay RY, is the main power relay. When the "Filament On" switch on the amplifier is thrown, the bias supply ( -150 volts) is energized. Power is applied to relay RY1 through the overload relay contacts (RY_B) and the time delay relay, TD. For usual voice operation, the plate supply is left on at all times. Relay RY1 may be released by the overload relay RY., whose actuator coil is placed in series with the B -minus lead of the high voltage supply. The overload relay is adjusted to trip at a plate current of approximately 850 ma. Once the relay is tripped, it is reset by the auxiliary reset coil by momentarily throwing off the filament switch. Screen voltage of the 4CX1000A is obtained from the high voltage supply through www.americanradiohistory.com 656 H.F. Power Amplifiers an adjustable dropping resistor and is controlled by two OD3 regulator tubes. With this regulator and divider combination, the screen voltage is stabilized at 300 volts, yet a maximum of only 10 watts may be drawn from the supply. This protects the 4CX 1000A under any operating conditions, as the maximum screen dissipation rating is 12 watts. In the event the plate supply fails, the tube is protected from screen overload, as the screen voltage is also removed at the same time. In case of bias failure, the plate circuit relay is de- energized, as it receives power from the bias supply. The screen current of the 4CX1000A varies over a wide range, depending upon the tube operating conditions, and may approach Figure 54 HINGED FRONT PANEL REVEALS BIAS AND VOLTMETER ADJUSTMENTS The main panel is hinged at the lower edge, and is cut out to permit meter switch and band switch knobs to project through. Special dial plates are cut from lucite and engraved for the two controls. At left, the two control potentiometers are mounted below meter switch. T H E R A D I O moderate negative values if the tube is lightly loaded. It is convenient, therefore, to be able to monitor negative values of screen current. A bleeder resistor is placed directly at the tube socket after the screen meter shunt. With 300 volts applied to the screen, this resistor is adjusted to provide a static reading of 20 milliamperes on the meter. Thus, 20 ma. must be subtracted from the meter reading to obtain the actual screen current. When the screen current is negative, the meter reading will drop below 20 milliamperes. A negative screen current of 18 ma., for example, will result in an indication of plus 2 ma. on the meter. Negative screen currents as great as -20 ma. can be monitored in this manner. The amplifier is cut off during standby periods by means of relay RY3, which boosts the grid bias to -150 volts. This prevents the amplifier from generating troublesome diode noise during periods of reception. Full operating potentials are applied to the amplifier at all times, and the amplifier is activated by removing the blocking bias. Relay RY3 may be controlled by an external voice circuit; it is only necessary to ground pin #2 to pin #3 on the control strip (figure 55) to activate the www.americanradiohistory.com HANDBOOK C.C. Ow--471---1 ORY,eI I 0 ' 42 01 - O 250 3000 V. I 100 SUPPLY I 115 V I I W 300 SEE TEXT RYI ADJUST V. V. OD3 OVERLOAD RY2e 0+3000 251c EA 200w 42301 A 657 RYIA 20A. 115V 4CX1000A Amplifier RY2 RESET COIL RY2 ,0 25w OD3 TO A.C. NEUT. TO A.C. NEUT. SEE TEXT 20 w I RY2A TI r7, SRI 1500 IÓ Ww -150V. RY3 SEE TEXT TO M.V. RECTIFIER FIL, AROMA - RY3 1ón 5w SRz - e 250 V. CONT O 2 BIAS J A.C. FIL. NEUT. SW. O 4 e+ O 115V. ON e+ SCR. e- e STRIPO 2 3 GND. TO CONTROL SW., ANT. RELAY, ETC. Figure 55 SCHEMATIC, AMPLIFIER CONTROL CIRCUIT RY1- Primary control relay. 115 volt a.c. coil or d.c. coil chosen to work with bias supply voltage. 13 ampere contacts. Potter it Brumfield type PR7AY RY3- Overload relay, l IS volt reset coil. Potter 8 Brumfield type GCIIA RY3 -SPST, 115 volt a.c. relay. Potter B Brumfield type KLSA SR1,2 -500 ma. rectifier. Sarkes- Tarzian M -500 T1 -150- 160 -170 volts @ 0.S amp. Triad R -93A Set at 170 volt tap TO- Thermal delay unit. Amperite 115 -NO -180 amplifier. The coil of the antenna relay may be placed in parallel with that of relay RY3 for completely automatic voice operation. Amplifier This amplifier is an excellent example of high -grade amateur construction. The unit is housed within an aluminum cabinet measuring 10" high, 15" wide and 151/2" deep. A false front panel, hinged at the lower edge (figure 54) is employed for decorative purposes. An auxiliary panel is placed behind this, holding the panel meters, control switches and the counter dials (figure 56). This auxiliary panel is spaced in front of the amplifier enclosure (figure 57) . Electrical connections between the amplifier and the auxiliary panel equipConstruction ment are made by means of two disconnect plugs, permitting the auxiliary panel to be wired and tested as a complete sub -assembly, or to be removed for servicing. Placement of the major components within the amplifier box may be seen in the top view, figure 58. No chassis is used, other than two shield boxes which enclose the tube socket and the power receptacles on the rear of the cabinet. The 4CX1000A tube is mounted on the top of an aluminum box measuring 6" square and 4" high. Only four connections pass into this compartment: Filament, screen, and bias leads (via coaxial capacitors seen in figure 57); and the excitation lead (via a coaxial plug and receptacle placed beneath the blower motor). www.americanradiohistory.com 658 H.F. Power Amplifiers Power and control leads from the shielded receptacle at the rear of the cabinet pass through a 3/4" aluminum conduit tube to the various circuitry mounted on the auxiliary front panel. The high voltage lead leaves the shield box via a short length of copper tubing to the bottom of the plate r.f. choke, which is supported from the rear wall of the cabinet. The variable vacuum capacitors are mounted to the tube socket assembly box by means of a heavy aluminum bracket, and are driven by the counter dials through flexible shaft couplers. The space between the front panel and the enclosure is about 31/4" and the filament transformer is mounted in the lower right portion of this area (figure 57). The enclosure is sealed by a hinged lid, which is TVI- proofed by phosphor- bronze finger stock fastened around the edge of the cabinet opening. The passive grid resistor (R1) is made up of nine 470 -ohm, 2 -watt composition resistors Figure 56 REAR VIEW OF METER PANEL Counter dial mechanisms, pilot lamps, meters and switches are mounted on aluminum sub panel. Meter switch, potentiometers, and meter resistors are mounted on phenolic panel at lower right. Panels have disconnect plugs so that they may be wired separately. T H E R A D I O placed in parallel (figure 59) . The resistor leads are clipped short, and the units are mounted between two copper discs, 11/4" in diameter. The assembly is enclosed in a copper tube measuring 11/2" inside diameter, 21/2" long, and having a 1/16" wall. After the resistor assembly is completed, it is bolted to a plate which fastens to one end of the tube. The container is then filled with transformer oil through a vent hole in the top. When it is full, it is placed in a pan of water which is heated to the boiling point. The oil expands and drives the air out through the second vent hole. Before the unit cools, the vent holes are closed with solder. This compact assembly will handle up to nearly 100 watts on an intermittent basis. If it is desired to make a less complex resistor assembly, thirty 1500 -ohm, 2 -watt composition resistors may be connected in parallel between two copper plates, in the manner shown in the photograph. This arrangement is cooled by the flow of air past the resistors. Amplifier Adjustment Before power is applied and Tuning to the amplifier, filament voltage should be adjusted to provide 6.0 volts at the socket of the 4CX 1000A. Voltage should be held within plus -or -minus 5 percent for maximum tube life, so an accurate voltmeter is required for this check. www.americanradiohistory.com HANDBOOK 4CX 1000A Amplifier 659 Figure 57 MAIN PANEL OF AMPLIFIER Meter multiplier and filament transformer (right) are mounted to the main panel. At left are feedthrough capacitors for various supply leads. Disconnect plugs to auxiliary panel are at center. Bandswitch shalt bracket is mounted to top of transformer. The amplifier is attached to a dummy antenna or other r.f. load. The sideband exciter may now be tuned and loaded, using the passive input resistor of the amplifier as a dummy load. The filament of the 4CX1000A is turned off, and high voltage applied to the amplifier. The reading of the screen current meter is noted, and the high voltage turned off and the screen bleeder adjusted until the meter indicates 10 milliamperes of bleeder current. The filament is now turned on, and the plate voltage applied and checked. The "Adjust Bias" potentiometer at the rear of the amplifier is set to provide a static plate current reading of 0.3 ampere. (Note that 60 ma. of indicated meter reading is current drawn by the screen regulator tubes and bleeder. This current is constant, regardless of plate current, and must be subtracted from the meter reading to obtain true plate current.) Next, apply a small carrier signal to the amplifier to increase the plate current indication by about 50 ma. A large value of negative screen current will be noted. Quickly set the loading capacitor to full scale, and adjust the plate tank capacitor for plate current dip, which will be very small. Monitor the screen current and advance the grid drive until about plus 10 to 20 milliamperes of screen current are noted. Decrease the capacitance of the loading capacitor (increase loading) slowly, and observe that the screen current decreases as the loading in- www.americanradiohistory.com 660 TH H.F. Power Amplifiers creases. Screen current will approach zero, and perhaps go slightly negative. Re- resonate plate tank, and increase excitation until plate current reaches 0.75 ampere. Screen current can be adjusted by alternately varying grid drive Figure 58 TOP VIEW OF AMPLIFIER CABINET Placement of the major components may be seen in this view. At left is 4CX1000A tube and socket, with blower immediately behind it. Atop the blower are the plate circuit ri. choke and blocking capacitors. Ten meter section of tank coil is mounted in a vertical position behind vacuum capacitor, which in turn is mounted to tube enclosure. At right is the tank inductor, with an auxiliary switch deck (not used) mounted on rear of assembly. This deck may be employed to switch antenna relays. Lid of cabinet is perforated to provide ventilation. Air intake is on left side of the cabinet beside blower cage. E RADIO and antenna loading. The sequence of events is to tune, load, and vary the drive until a plate meter reading of 0.75 ampere is achieved, with an indicated screen current of approximately 0 to plus 20 ma. When excitation is removed, screen current will drop to about 18 ma. This indicates a true resting screen ma., plus a bleeder current of current of 20 ma. Grid current, of course, is zero. The carrier signal may now be removed, and voice excitation applied to the amplifier. Plate current may be "talked" up to about 0.38 ampere on voice peaks. True screen current will run -5 ma. to plus 20 ma. on voice peaks, depending upon the degree of loading and the exact ratio between loading and grid drive. Under optimum conditions, screen current will -2 www.americanradiohistory.com HANDBOOK 3 -1000Z rest at 8 ma., and drop to about voice waveforms. 29 -13 -2 ma. under A 3 -1000Z Linear Amplifier The 3 -1000Z is a compact power triode intended to be used as a zero bias class B amplifier in audio or r.f. applications. Grounded grid linear service is especially attractive, as full legal input may be run at a plate potential of only 2500 volts, yet the power gain of the tube is high enough to allow sideband exciters of the "100 watt" -type to drive it to full output. Neutralization of the grounded grid stage is not necessary, as the excellent internal shielding of the tube reduces intra -stage feedback to a minimum. Distortion products of this amplifier are better than 35 decibels below maximum p.e.p. level. A tuned cathode tank circuit is employed in order to obtain greatest linearity and power output. Amplifier Circuit The 3 -1000Z amplifier covers all amateur bands between 3.5 Mc. and 29.7 Mc. with generous overlaps. Bandswitching circuits are used, and the unit is designed to operate with unbalanced coaxial antenna systems having an s.w.r. of less than 2/1. The complete schematic is given 661 in figure 61. A high -C, bandswitching cathode circuit is used for best linearity (figure 62) . The driving impedance of the 3 -1000Z is approximately 55 ohms, providing a close match to the great majority of sideband exciters. The "flywheel effect" of the cathode tank prevents input waveform distortion caused by the halfcycle loading of the class B amplifier. Filament voltage is fed to the tube via a shunt choke (L2) placed in parallel with the tuned circuit. The cathode coil is tapped for the various amateur bands, and extra shunt capacity is placed in the circuit to maintain the proper C/L ratio at 3.5 Mc. Plate current metering is accomplished in the B -minus lead to the power supply to remove dangerous voltages from the meter movement. The meter is shunted with a wirewound resistor as a safety measure. For standby operation, the cathode to ground return of the stage is opened by means of the voice relay, and a small amount of idling current flowing through a 50K cathode resistor provides sufficient bias to prevent annoying diode noise from being generated during listening periods. The voice relay shorts out the resistor to allow normal operation of the stage. It is necessary to "unground" the grounded grid sufficiently to permit measurements of grid current to be made. The 3 -1000Z has ii 1( Figure 59 NONINDUCTIVE 50 OHM GRID Nine Amplifier RESISTOR composition 2 -watt resistors are immersed in oil bath to provide high peak level of dissipation. Exciter may be tuned up using this resistor as a dummy load, if desired. www.americanradiohistory.com H.F. Power Amplifiers 662 three grid pins, and each corresponding socket terminal is bypassed to ground with a low impedance r.f. shunt made of a 4.7 -ohm composition resistor and a 0.01 pfd., 1.2 kv. ceramic disc capacitor connected in parallel. Figure 60 2 KILOWATT P.E.P. GROUNDED - GRID LINEAR AMPLIFIER 3 -10002 zero bias triode tube is designed for grounded -grid linear amplifier service, and is capable of full input at a plate potential as low as 2500 volts. This 3 -1000Z amplifier covers all amateur bands The Eimac between 10 and 80 meters. Panel meters and controls are (I. to r.): plate, Arid and output meter; plate tuning (center); bandswitch; cathode bandswitch and tuning (lower left); antenna loading (center) and output voltmeter adjustment. Complete amplifier is enclosed in screen made of perforated Reynolds aluminum sheet. The voltage drop across the resulting resistance (1.6 ohms) is measured by a simple d.c. voltmeter made up of a 0 -1 d.c. milliammeter with a series multiplier chosen to provide a full scale reading when 300 ma. of grid current develops 0.64 volts across the shunt. The internal resistance of the meter is subtracted from the value of the required series multiplier. A pi- network tank circuit is used, with an additional loading capacitor that can be switched in the circuit to match low impedance antenna loads commonly encountered on the 80 meter band. In addition, a diode voltmeter is included to monitor the output voltage of the amplifier. The 3 -1000Z requires forced -air cooling to maintain the base seals at a temperature below 200 °C, and the plate seal at a temperature www.americanradiohistory.com a+OOH N N Q ÿdo-n úáo É C tt iW 213, 13, 0,0 0N Ó Old O 7,-t-4 n cEEE "If.;0 Etóf bEoó ° E ñ 3 E W à ç.2 m°Eâo>°d. u ut" 1G ug ... OX O ^ O` Ç d d 3 d O;] t O O w~ H G « 0 O d E«d OI C7 ó u 7 d 4. mL L d ú úO C ó lptE110276 0+ X} Oa 2,'T RI° ---+w--1n á ry J_ óIN SU ó o tó C P., r~4 US 12 UR a 2 r va u. z 00 rW ad 13 o c v o o Ñ ñ o n Ó o d ' 3 a ,. Oi a o N O 7 7 _ Fl é O o et N « o r u << 44 -I. -0, - N d o a p ,n 0 a 0 0 W Q a W ó«3 d d d um C u` a ÿx ' á w G Y óo i et Od - ° 0ám G a- u «Y ; « ^If O Z G °0 M 0 y «C OpQ`LPF-2- LOW CHIC,AISGO RC -12130 2A. DAG SR -50 MA. REPLACEMENT TYPE SELENIUM RECTIFIER IS RESISTORS MARKED WITH ASTERISK K PA /RS. MAX. ADJUST CLIPPING T1 CLASS MODULATOR GRID CIRCUIT UNLESS OTHERWISE INDICATED; ALL RESISTORS 0.3 WATT ALL CAPACITORS IN LF 1 2 Ti 50 1 W. SR TD 1113V ,N ONO. 0- Figure 13 SCHEMATIC, 15 WATT CLIPPER -AMPLIFIER A three wire shielded cable should be used to connect the 6B4 -G tubes to the driver transformer located at the grids of the class B tubes. This cable may be any reasonable length up to 25 or 30 feet. Any of the modulator configurations shown in figure 8 may be driven with this simple speech amplifier. 30 -5 A 15 -Watt Clipper - Amplifier The near -ultimate in 'talk power" can be obtained with low level clipping and filtering combined with high level filtering. Such a modulation system will have real "punch," yet will sound well rounded and normal. The speech amplifier described in this section makes use of low level clipping and filtering and is specifically designed to drive a pair of push -pull 810 modulators such as shown in Section Three. ping control, R2. Amplifier gain is controlled by R,, in the grid circuit of the second section of the 12AX7. A low pass filter having a 3500 cycle cut -off follows the 6AL5 clipper stage, with an output of 5 volts peak audio signal under maximum clipping conditions. A double triode 12AU7 cathode follower phase- inverter follows the clipper stage and delivers a 125 volt r.m.s. signal to the push -pull grids of the 6B4 -G audio driver tubes. The 6B4 -G tubes operate at a plate potential of 330 volts and have a -68 volt bias voltage developed by a small selenium rectifier supply applied to their grid circuit. An audio output of 15 watts is developed across the secondary terminals of the class B driver transformer with less than 5 per cent distortion under conditions of no clipping. A 5U4 -G and a choke input filter network provide unusually good voltage regulation of the high voltage plate supply. Circuit Amplifier Description Construction The schematic of the speech amplifier -clipper is shown in figure 13. A total of six tubes, including a rectifier are employed and the unit delivers 15 watts of heavily clipped audio. A 12AX7 tube is used as a two stage microphone pre-amplifier and delivers approximately 20 volts (r.m.s.) audio signal to the 6AL5 series clipper tube. The clipping level is adjustable between 0 db and 15 db by clip- The clipper-amplifier may be built upon the same chassis as the power supply, provided the low level stages of the amplifier are spaced away from the power transformers and filter chokes of the supply. All capacitors and resistors of the audio section should be mounted as close to the respective sockets as is practical. For minimum hum pickup, the filament leads to the low level stages should be run in shielded braid. www.americanradiohistory.com HANDBOOK 15 -Watt Those resistors in the 12AU7 phase inverter plate circuit and the grid circuit of the 6B4 -G tubes should be matched to achieve best phase inverter balance. The exact value of the paired resistors is not important, but care should be taken that the values are equal. Random resistors may be matched on an ohmmeter to find two units that are alike in value. When these matched resistors are soldered in the circuit, care should be taken that the heat of the soldering iron does not cause the resistors to shift value. The resistors should be held firmly by the lead to be soldered with a long nose pliers, which will act as a heat -sink between the soldered joint and the body of the resistor. If this precaution is taken the two phase inverter outputs will be in close balance. When the wiring of the speech amplifier has been completed and checked, the unit is ready to be tested. Before the tubes are plugged in the amplifier, the bias supply should be energized and the voltage across the 600 ohm bleeder resistor should be measured. It should be -68 volts. Adjustment of the Speech Amplifier If it is not, slight changes in the value of the series resistor, Rr, should be made until the correct voltage appears across the bleeder resistor. The tubes may now be inserted in the amplifier and the positive and cathode voltages checked in accordance with the measurements given in figure 13. After the unit has been Clipper -Amplifier 679 tested and is connected with the modulator, R2 should be set so that it is impossible to over modulate the transmitter regardless of the setting of R1. The gain control (R.) may then be adjusted to provide the desired level of clipping consistent with the setting of R2. 30 -6 A 200 -Watt 811 -A De -luxe Modulator One of the most popular medium power r -f amplifier stages consists of a single tetrode tube, such as the 4 -125A, 813, or 7094 operating at a plate potential of 1200 - 1700 volts and a plate current of 150 - 275 milliamperes. Such an amplifier requires a minimum of r -f driving power, allows an input of 300 to 400 watts, and yet employs power supply components that are relatively modest in cost. The 5 -db signal increase between a 300 watt transmitter and a 1000 watt transmitter is very expensive when one considers the additional cost of modulator and power supply equipment. Additional economy may be achieved if the modulator and final amplifier are operated from the same power supply. The new series of Chicago- Standard plate transformers provide voltage ranges in the 1000 to 1500 volt region and are well suited for the combination of this modulator and the aforementioned r -f tubes. Within this voltage range, the 811 -A triode is an excellent choice for the modulator tubes. Zero bias operation may be had up to Figure 14 REAR VIEW OF 811 -A MODULATOR Modulator tubes and voltage regulator are at right with high level filter at center of chassis. Plug -in speech amplifier is to left of clipper. Gain and clipping controls are atop the small chassis. 6L6'5881 is used as cathode follower driver stage for modulator. www.americanradiohistory.com THE RADIO Speech and A. M. Equipment 680 PLUG -IN SPEECH AMPLIFIER CRYSTAL MIC 503 6AL5 12AX7 -- S01!PLI SN v Ti 12AU7 L1 2 1M SK 10UF - 1W 25 + 1.6 TE6.362v L J 611F- 61JF450 450 - v N Dc1_ PL2,, L NOTES I 2 SO4 Y -ALL RESISTORS 1/Z WATT UNLESS 0ÚF OTHERWISE SPECIFIED. -SW ITCH Si (PHONE-CW) SHOWN IN PHONE POSITION. 1 6L6GB /5881 3- RY1 SHOWN ENERGIZED. 12 6 6L6GB OF ISO 10 AND 2D21 FILS 2 _J SOS _ 3 0 4W 3 2- j63 ti V. OUT 3- GROUND. 6'- _ / /3V ti /N 6- 8#330 V TO EXCITER 7 -1 CW -PHONE SWITCH 6- I 9- GROUND, B1011- RY1 CONTROL 12-MIC. CONTROL PUSH -TO-TALK CIRCUIT 7 e 811 -A 004 3NV 10 +HV OUr TO AMPLIFIER +HV IN A 811 TS RY1 2D21 Figure 15 SCHEMATIC, 200 WATT 811 -A MODULATOR 1, -1:3 Interstage Transformer. Stanco, A -53 "Poly -pedante" class B driver transformer. 2:1 ratio. Stanco, A -4761 T,-200 watt modulation transformer. 9 K primary. 5K secondary. Stanco, A -3829 T.-400 - 0 - 400 volts, 250 ma., 6.3 volts, 5 amperes. Stanco, PC -8413 T,-6.3 volts, 10 amperes. Stanco, P -6308 CH,-4 henry, 250 ma. Stanco, C -1412 L,-Low pass filter, 3000 cycle cut off. Chicago LPF -2. "Splatter" filter, 300 ma. Stanco, C -2317 RY,-SPDT relay, high voltage insulation, 115 volt coil. Leach =1723 with 374 coils, or equivalent Tr- L,- 1250 volts, and only -4.5 volts is required for 1500 volt operation. Bias voltage may be obtained from flashlight batteries or other low impedance source. The 200 watt de -luxe modulator is illustrated in figures 14 and 16 and the schematic is given in figure 15. The low level audio stages are similar to those of the speech amplifier shown in Section Six. A 12AX7 is employed as two Modulator Circuit stages of R -C amplification driving a 6AL5 speech clipper tube. A 3500 cycle low pass filter follows the clipper, removing all high order products of clipping action. A parallel connected 12AU7 follows the filter and is transformer -coupled to a 5881 (6L6 -GB) cathode follower driver stage. The impedance of the cathode circuit of the driver stage is extremely low; it provides an excellent driving source for the class B modulator grid circuit. www.americanradiohistory.com HANDBOOK 81 Two 811 -A tubes are employed in the class B stage. When operated at 1000 volts, no bias supply is needed. At voltages of 1200 or above, approximately 9 volts of bias is required. This is supplied by a voltage divider composed of a 20K, 10 watt resistor and a 2D21 thyratron tube. When the miniature 2D21 is connected as a triode, it acts as a voltage regulator tube with a constant voltage drop of almost 9 volts from plate to cathode. The tube will regulate over 300 milliamperes of current while maintaining a reasonably constant voltage drop across its terminals. The center tap of the 811 -A filament transformer (Ta) is thus held at a positive potential with respect to ground. Since the center tap of the 811 -A driver transformer (T2) is grounded, the modulator tubes are biased at a constant negative voltage equal to the voltage drop across the 2D21 regulator tube in the cathode circuit of the class B stage. The plate to plate load impedance of the 811 -A tubes when operating at 1500 volts is approximately 12,000 ohms. A multi -match type modulation transformer may be employed if desired, but in this case a Stancor A -3829 unit was used. This transformer is designed to match the plate -to -plate load impedance of 9,000 ohms to a secondary load of 5000 ohms. With the 12,000 ohm load of the 811 -A tubes, a secondary load of 7,500 ohms must be used to maintain the same primary to secondary impedance ratio. This secondary load can be obtained with a single 7094 tube operating at 1500 volts and 200 milliamperes of plate current (300 watts input) . Other tubes and load impedances can also be used, providing the r -f input to the modulated stage does not exceed 400 watts. For example, a 4 -125A tube operating at 2000 volts and 165 ma. (330 watts) may be modulated by this audio unit. The secondary winding of the modulation transformer can pass a maximum of 300 milliamperes with safety. The audio output from the 811 -A stage is passed through a high level low -pass "splatter suppressor" which attenuates all audio frequencies above 3500 cycles. The use of both low level and high level audio filters does 1 -A Modulator 681 much to reduce the broad sidebands and cochannel interference that seems to be so common on the amateur phone bands. A high voltage relay RN', is employed to short the secondary of the modulation transformer and remove plate potential from the modulator tubes for c -w operation. The relay is actuated by the "phone -c.w." switch on the front panel of the modulator. Other segments of this switch turn off the modulator filaments and provide extra contacts to control auxiliary equipment. A 350 volt supply is incorporated in the modulator unit to power the speech amplifier and driver stage and to provide power for the r -f exciter stages of the transmitter. The various power and control leads are brought out to a multi -connector plug mounted on the rear of the modulator deck. Figure 16 UNDER -CHASSIS VIEW OF 811 -A MODULATOR High voltage relay is between 811 -A tube sockets, and low voltage components ore at opposite end of chassis. www.americanradiohistory.com 682 Speech and A. M. Equipment Modulator Construction The modulator is constructed upon a steel chassis measur".A 101/2" ing 8" x 17 aluminum panel is bolted to the chassis with two mounting brackets to form a rugged assembly. Placement of the major parts may be seen in figures 14 and 16. The modulation transformer T3 and the 811 -A tubes occupy the right end of the chassis, balanced in weight by the power transformer T. and modulator filament transformer T. at the opposite end of the chassis. The center space is taken by the plug -in speech amplifier, the high level splatter filter assembly and the 5881 driver stage. The speech amplifier is constructed as a separate unit on a small aluminum utility box measuring 5" x 3" x 2 ". The bottom of the "x2 box holds two male plugs which match two receptacles mounted on the amplifier chassis. The speech amplifier, therefore, may be wired and tested as a separate unit. Clipping and audio level controls are mounted atop the amplifier box as long usage of clipper circuits has proven that these controls need not be readjusted once they are properly set. The phone -c.w. switch, relay RY-1 and various small components are mounted beneath the chassis (figure 16) . The input receptacle for the speech amplifier box is located adjacent to the microphone receptacle on the front panel of the modulator making the interconnecting lead less than two inches long. Also placed beneath the chassis are the filter choke for the low voltage supply and the various bypass and filter capacitors. Wiring and Testing the Modulator The speech amplifier should be wired first. The small resistors and capacitors are mounted either between the tube socket pins, or between terminals of small phenolic tie -point strips. Transformer T. is fastened within the amplifier box and is wired in the circuit after all other wiring is completed. Plugs PL, and PL: are mounted on the bottom portion of the box; the plug pins are wired to the proper points of the speech amplifier with short lengths of wire that allow the bottom plate to be removed for inspection and testing without the necessity of unsoldering any connections to the plugs. The modulator chassis should be wired next. All leads to T3, RY -3, and the low pass filter should be carefully insulated from the chassis. High voltage "5000 volt test" cable should be employed for these connections. The capacitors that make up the high level audio filter are mounted directly to the terminals of the T H E R A D I O filter choke which is mounted above the chassis on 1/2-inch ceramic insulators. High voltage connections to the modulator are made through Millen 37001 safety terminals. When the wiring has been completed and checked, the 12AX7, 6AL5, 12ÁU7, 5881, and 5V4 -GB tubes should be inserted in their sockets and the speech amplifier is plugged into the modulator receptacles. The vertical amplifier of an oscilloscope should be connected to one grid terminal of the 811 -A stage. Plate voltage of the 5881 should be approximately 370 volts. A low level 1000 cycle tone (approximately 0.05 volts, r.m.s.) is applied to the amplifier input. The output level of the speech amplifier is controlled by the setting of the clipping control Rz and the audio gain is controlled by potentiometer R1 in the grid circuit of the 12AX7. The clipping control should be set so that not more than 60 volts r.m.s. is developed from one 811 -A grid to ground. The modulator tubes may now be plugged in their sockets. A 7K, 200 watt resistor should be placed between the "H.V. Out" and " H.V. In" terminals, serving as a dummy load, and 1500 volts applied to the latter terminal. With no audio signal the resting plate current of the modulator stage should be approximately 15 milliamperes, kicking up to about 160 milliamperes under full output conditions. Final adjustment of the clipping control may be made when the modulator is placed in use with the r -f section of the transmitter. Potentiometer Rz is then adjusted to limit the peak modulation level under sine wave modulation to approximately 90%. V+ TO DRi VER STAGE Figure 17 ZERO BIAS TETRODE MODULATOR ELIMINATES SCREEN AND BIAS SUPPLIES driving power and simplicity are key features of this novel modulator. Tubes ranging in size from 6AQ5's to 813's may be employed in this circuit. Low T -Class 8 driver transformer Tr- Modulation -6AQ5, V,, V. R, R: -Not transformer 6L6, 807, 803, 813, etc. used with 803 and 813 www.americanradiohistory.com HANDBOOK 683 803's T 6 -WATT SPEECH AMP. (F /G /2) 2 1 II : _ /70 DRIVING POWER= TO MODULATOR LOAD - OPERATING CHARACTERISTICS EGG (RMS) ZS =6 25K - 8+2500 V. VOLTS 7 -6 WATTS RESTING PLATE CUR.= SOMA. MAX. PLATE CUR =340 MA. POWER OUTPUT = 5/0 SUPPRESSOR VOLT = WATTS 260 -340v IIS U 1 Figure 18 INEXPENSIVE 500 WATT MODULATOR USING 803 TUBES T,- "Poly -pedance" Class B driver transformer 2:1 ratio. Stancor A -4761 18K primary, 6.25K secondary. Chicago CMS -3 Tr-500 watt output transformer. T, -10 volts, 10 amperes. M-0 - 500 mo. Stancor C -6461 Zero Bias Tetrode Modulators 30 -7 Class B zero bias operation of tetrode tubes is made possible by the application of the driving signal to the two grids of the tubes as shown in figure 17. Tubes such as the 6AQ5, 6L6, 807, 803, and 813 work well in this circuit and neither a screen supply nor a bias supply is required. The drive requirements are low and the tubes operate with excellent plate circuit efficiency. The series grid resistors for the small tubes are required to balance the current drawn by the two grids, but are not needed in the case of the 803 and 813 tubes. Of great interest to the amateur is the circuit of figure 18, wherein 803 tubes are used as high level modulators. These tubes will deliver 500 watts of audio in this configuration, yet they require no screen or bias supply, and can be driven by an 8 watt amplifier stage. The use of 803 tubes (in contrast to 813's) requires a higher level of driving power which is offset by the fact that these tubes can often be purchased "surplus" for less than four dollars. A pair of 6B4 tubes operating with cathode bias (figure 12) will suffice as a driving stage for the 803's. The power supply of the speech amplifier provides high voltage for the suppressors of the modulator stage. Shown in figure 19 is a high level modulator using 813 tubes. A full 500 watts of audio may be obtained at 2500 volts plate potential. Grid driving power is 5.5 watts. A single 807 operating as a cathode follower at 400 volts will provide sufficient drive for the modulator stage. Plate to plate load impedance for the 813's is not critical. The Chicago CMS -3 500 watt modulation transformer having a primary impedance of 18,000 ohms has been used with success, although the optimum plate load impedance of the modulator is closer to 20,000 ohms. 807 6C4,6J5 Erc. T 813,5 TO MODULATOR LOAD +400 DRIVER STAGE,OPERATING VOLTS, MEASURED TO GROUND. PIN 2 807 PIN 300 .3 PIN 26,5 NOTES I X X -EXACT VALUE OF 607 CATHODE RESISTOR DEPENDS UPON RESISTANCE OF PRIMA RV WINDING OF T2. ADJUST RESISTOR FOR CATHODE 8 /AS OF 26.5 VOL rs, PLATE CURRENT OF 53 -55 MA. 2 -RMS OPERATING VOLTAGES AT MAXIMUM OUTPUT SHOWN ON SCHEMATIC. 111 11SV.rt. Figure 19 500 WATT MODULATOR USING 813 TUBES T,-1:3 interstage transformer. Stancor A -53 Tr- "Poly -pedance" Class B driver transformer. T,-500 watt output transformer. 18K primary, T.-10 volts, 10 amperes, Stancor C-6461 2:1 ratio. Stancor A -4761 6.25K secondary. Chicago CMS -3 M-0 - 500 ma. 350 watts of audio are obtainable from this circuit at plate potential of 2000 volts. www.americanradiohistory.com CHAPTER THIRTY -ON E In view of the high cost of iron -core components such as go to make up the bulk of a power supply, it is well to consider carefully the design of a new or rebuilt transmitter in terms of the minimum power supply requirements which will permit the desired performance to be obtained from the transmitter. Careful evaluation of the power supply requirements of alternative transmitter arrangements will permit the selection of that transmitter arrangement which requires the minimum of power supply components, and which makes most efficient use of such power supplies as are required. 31 -1 Power Supply Requirements A power supply for a transmitter or for a unit of station equipment should be designed in such a manner that it is capable of delivering the required current at a specified voltage, that it has a degree of regulation consistent with the requirements of the application, that its ripple level at full current is sufficiently low for the load which will be fed, that its internal impedance is sufficiently low for the job, and that none of the components shall be overloaded with the type of operation contemplated. The meeting of all the requirements of the previous paragraph is not always a straightforward and simple problem. In many cases compromises will be involved, particularly when the power supply is for an amateur station and a number of components already on hand must be fitted into the plan. As much thought and planning should be devoted to the power- supply complement of an amateur station as usually is allocated to the r -f and a -f components of the station. The arrival at the design for the power supply for use in a particular application may best be accomplished through the use of a series of steps, with reference to the data in this chapter by determining the values of components to be used. The first step is to establish the operating requirements of the power supply. In general these are: 1. Output voltage required under full load. 2. Minimum, normal, and peak output current. 3. Voltage regulation required over the current range. 684 www.americanradiohistory.com Requirements Figure 1 POWER SUPPLY FILAMENT CONTROL PANEL VOLTAGE PLATE VOLTAGE designed supply conhas separate primary switches and indicator lamps for the filament and plate circuits, overload circuit breaker, plate voltage control switch and primary circuit fuses. A 685 well trol panel 0 3500 411 PLATE VOLTAGE FINAL BIAS 2 AMP 2 AMP SCREEN 2 AMP. MN 4. Ripple voltage limit. 5. Rectifier circuit to be used. The output voltage required of the power supply is more or less established by the operating conditions of the tubes which it will supply. The current rating of the supply, however, is not necessarily tied down by a particular tube combination. It is always best to design a power supply in such a manner that it will have the greatest degree of flexibility; this procedure will in many cases allow an existing power supply to be used without change as a portion of a new transmitter or other item of station equipment. So the current rating of a new power supply should be established by taking into consideration not only the requirements of the tubes which it immediately will feed, but also with full consideration of the best matching of power supply components in the most economical current range which still will meet the requirements. It is often long run economy, however, to allow for any likely additional equipment to be added in the near future. Current- Rating Considerations The minimum current drain which will be taken from a power supply will be, in most cases, merely the bleeder current. There are many cases where a particular power supply will always be used with a moderate or heavy load upon it, but when the supply is a portion of a transmitter it is best to consider the mini- MOD 2AMMP. EMISSION FONECA \ MN mum drain as that of the bleeder. The minimum current drain from a power supply is of importance since it, in conjunction with the nominal voltage of the supply, determines the minimum value of inductance which the input choke must have to keep the voltage from soaring when the external load is removed. The normal current rating of a power supply usually is a round -number value chosen on the basis of the transformers and chokes on hand or available from the catalog of a reliable manufacturer. The current rating of a supply to feed a steady load such as a receiver, a speech amplifier, or a continuously-operating r -f stage should be at least equal to the steady drain of the load. However, other considerations come into play in choosing the current rating for a keyed amplifier, an amplifier of SSB signals, or a class B modulator. In the case of a supply which will feed an intermittent load such as these, the current ratings of the transformers and chokes may be lets than the maximum current which will be taken; but the current ratings of the rectifier tubes to be used should be at least equal to the maximum current which will be taken. That is to say that 300 -ma. transformers and chokes may be used in the supply for a modulator whose resting current is 100 ma. but whose maximum current at peak signal will rise to 500 ma. However, the rectifier tubes should be capable of handling the full 500 ma. www.americanradiohistory.com 686 T H Power Supplies The iron -core components of a power supply which feeds an intermittent load may be chosen on the basis of the current as averaged over a period of several minutes, since it is heating effect of the current which is of greatest importance in establishing the ratings of such components. Since iron -core components have a relatively large amount of thermal inertia, the effect of an intermittent heavy current is offset to an extent by a key -up period or a period of low modulation in the case of a modulator. However, the current rating of a rectifier tube is established by the magnitude of the emission available from the filament of the tube; the maximum emission must not be exceeded even for a short period or the rectifier tube will be damaged. The above considerations are predicated, however, on the assumption that none of the iron -core components will become saturated due to the high intermittent current drain. If good -quality components of generous weight are chosen, saturation will not be encountered. Voltage Regulation The general subject of voltage regulation can really be divided into two sub -problems, which differ greatly in degree. The first, and more common, problem is the case of the normal power supply for a transmitter modulator, where the current drain from the supply may vary over a ratio of four or five to one. In this case we desire to keep the voltage change under this varying load to a matter of 10 or 15 per cent of the operating voltage under full load. This is a quite different problem from the design of a power supply to deliver some voltage in the vicinity of 250 volts to an oscillator which requires two or three milliamperes of plate current; but in this latter case the voltage delivered to the oscillator must be constant within a few volts with small variations in oscillator current and with large variations in the a -c line voltage which feeds the oscillator power supply. An additional voltage regulation problem, intermediate in degree between the other two, is the case where a load must be fed with 10 to 100 watts of power at a voltage below 500 volts, and still the voltage variation with changes in load and changes in a -c line voltage must be held to a few volts at the output terminals. These three problems are solved in the normal type of installation in quite different manners. The high -power case where output voltage must be held to within 10 to 15 per cent is normally solved by using the proper value of inductance for the input choke and proper E R A D I O value of bleeder at the output of the power supply. The calculations are simple: the inductance of the power-supply input choke at minimum current drain from the supply should be equal in henries to the load resistance on the supply (at minimum load current) divided by 1000. This value of inductance is called the critical inductance and it is the minimum value of inductance which will keep the output voltage from soaring in a choke -input power supply with minimum load upon the output. The minimum load current may be that due to the bleeder resistor alone, or it may be due to the bleeder plus the minimum drain of the modulator or amplifier to which the supply is connected. The low- voltage low- current supply, such as would be used for a v.f.o. or the high- frequency oscillator in a receiver, usually is regulated with the aid of glow- discharge gaseous regulator tubes. These regulators are usually called "VR tubes." Their use in various types of power supplies is discussed in Section 31 -9. The electronically -regulated power supply, such as is used in the 10 to 100 watts power output range, also is discussed later on in this chapter, The ripple -voltage limitation imposed upon a power supply is determined by the load which will be fed by the supply. The tolerable ripple voltage from a supply may vary from perhaps 5 per cent for a class B or class C amplifier which is to be used for a c -w stage or amplifier of an FM signal down to a few hundredths of one per cent for the plate-voltage supply to a low-level voltage amplifier in a speech amplifier. The usual value of ripple voltage which may be tolerated in the supply for the majority of stages of a phone transmitter is between 0.1 and 2.0 per cent. In general it may be stated that, with 60cycle line voltage and a single -phase rectifier circuit, a power supply for the usual stages in the amateur transmitter will be of the choke input type with a single pi- section filter following the input choke. A c -w amplifier or other stage which will tolerate up to 5 per cent ripple may be fed from a power supply whose filter consists merely of an adequate -size input choke and a single filter capacitor. A power supply with input choke and a Ripple Considerations single capacitor also will serve in most cases to feed a class B modulator, provided the output capacitor in the supply is sufficiently large. The output capacitor in this case must be capable of storing enough energy to supply the www.americanradiohistory.com HANDBOOK Requirements peak- current requirement of the class B tubes on modulation peaks. The output capacitor for such a supply normally should be between 4 pfd. and 20 pfd. Capacitances larger than 20 µfd. involve a high initial charging current when the supply is first turned on, so that an unusually large input choke should be used ahead of the capacitor to limit the peak- current surge through the rectifier tubes. A capacitance of less than 4 pfd. may reduce the power output capability of a class B modulator when it is passing the lower audio frequencies, and in addition may superimpose a low -frequency "growl" on the output signal. This growl will be apparent only when the supply is delivering a relatively high power output; it will not be present when modulation is at a low level. When a stage such as a low -level audio amplifier requires an extremely low value of ripple voltage, but when regulation is not of importance to the operation of the stage, the high degree of filtering usually is obtained through the use of a resistance- capacitance filter. This filter usually is employed in addition to the choke -capacitor filter in the power supply for the higher -level stages, but in some cases when the supply is to be used only to feed low- current stages the entire filter of the power supply will be of the resistance- capacitance type. Design data for resistance -capacitance filters is given in a following paragraph. When a low- current stage requires very low ripple in addition to excellent voltage regulation, the power supply filter often will end with one or more gaseous -type voltage- regulator tubes. These VR tubes give a high degree of filtering in addition to their voltage-regulating action, as is obvious from the fact that the tubes tend to hold the voltage drop across their elements to a very constant value regardless of the current passing through the tube. The VR tube is quite satisfactory for improving both the regulation and ripple characteristics of a supply when the current drain will not exceed 25 to 35 ma. depending upon the type of VR tube. Some types are rated at a maximum current drain of 30 ma. while others are capable of passing up to 40 ma. without damage. In any event the minimum current through the VR tube will occur when the associated circuit is taking maximum current. This minimum current requirement is 5 ma. for all types of gaseous -type voltage -regulator tubes. Other types of voltage -regulation systems, in addition to VR tubes, exhibit the added RIPPLE IN TERMS OF C AT FULL LOAD FULL- WAVE RECTIFIER TO 5 -25 NY 687 PERCENT RIPPLE 13.1 CAPACITANCE, C 2 UF 3 LF 25000 6.5 6.2 4.0 UF 6 L: FIGURE 2 TO RIPPLE IN TERMS OF LOAD RESISTANCE FULL-WAVE RECTIFIER PERCENT RIPPLE LOAD. ONMS 25000 (BLEEDER ONLY) 0.02 0.04 15000 25000 10000 0.06 5000 3000 2 000 0.1 O 17 0.25 FIGURE 3 TO FULL -WAVE RIPPLE IN TERMS OF CI AND C2 AT FULL LOAD RECTIFIER C1 C2 2 2 1.2 0.7 3 25000 PERCENT RIPPLE 4 a 0 B 6 0.06 25 FIGURE 4 characteristic of offering a low value of ripple across their output terminals. The electronic -type of voltage -regulated power supply is capable of delivering an extremely small value of ripple across its output terminals, even though the rectifier- filter system ahead of the regulator delivers a relatively high value of ripple, such as in the vicinity of 5 to 10 per cent. In fact, it is more or less self evident that the better the regulation of such a supply, the better will be its ripple characteristic. It must be remembered that the ripple output of a voltage -regulated power supply of any type will rise rapidly when the load upon the supply is so high that the regulator begins to lose control. This will occur in a supply of the electronic type when the voltage ahead of the series regulator tube falls below a value equal to the sum of the minimum drop across the tube at that value of current, plus the output voltage. In the case of a shunt regulator of the VR -tube type, the regulating effect will fail when the current through the VR tube falls below the usual minimum value of about 5 ma. Although figures 2, 3 and 4 give the value of ripple voltage for several more or less standard types of filter systems, it is often of value to be able to calculate the value of ripple voltage to be expected with a particular set of filter components. Fortunately, the approximate ripple percentage for normal values Calculation of Ripple www.americanradiohistory.com 688 Power Supplies e Hr T H 12 HT FULL-WAVE RECTIFIER TO Figure 5 SAMPLE FILTER FOR CALCULATION OF RIPPLE of filter components may be calculated with the aid of rather simple formulas. In the two formulas to follow it is assumed that the line frequency is 60 cycles and that a full wave or a full -wave bridge rectifier is being used. For the case of a single -section choke -input filter as illustrated in figure 2, or for the ripple at the output of the first section of a two -section choke input filter the equation is as follows, 118 Per cent ripple = LC -1 where LC is the product of the input choke inductance in henrys (at the operating current to be used) and the capacitance which follows this choke expressed in microfarads. In the case of a two -section filter, the per cent ripple at the output of the first section is determined by the above formula. Then this percentage is multiplied by the filter reduction factor of the following section of filter. This reduction factor is determined through the use of the following formula: LC -1 Filter reduction factor 1.76 Where LC again is the product of the inductance and capacitance of the filter section. The reduction factor will turn out to be a decimal value, which is then multiplied by the percentage ripple obtained from the use of the preceding formula. As an example, take the case of the filter diagrammed in figure 5. The LC product of the first section is 16. So the ripple to be expected at the output of the first section will be: 118/ (16 -1) or 118/15, which gives 7.87 per cent. Then the second section, with an LC product of 48, will give a reduction factor of: 1.76/ (48 -1) or 1.76/47 or 0.037. Then the ripple percentage at the output of the total filter will be: 7.87 times 0.037 or slightly greater than 0.29 per cent ripple. - Resistance- Capacitance Filters In many applications where the current drain is relatively small, so that the voltage drop across the series resistors would not be E R A D I O excessive, a filter system made up of resistors and capacitors only may be used to advantage. In the normal case, where the reactance of the shunting capacitor is very much smaller than the resistance of the load fed by the filter system, the ripple reduction per section is equal to 1/ (2TrRC). In terms of the 120 -cycle ripple from a full -wave rectifier the ripple -reduction factor becomes: 1.33 /RC where R is expressed in thousands of ohms and C in microfarads. For 60 -cycle ripple the expression is: 2.66/RC with R and C in the same quantities as above. Filter System Many persons have noticed, particularly when using an input choke followed by a 2 -µfd. first filter capacitor, that at some value of load current the power supply will begin to hum excessively and the rectifier tubes will tend to flicker or one tube will seem to take all the load while the other tube dims out. If the power supply is shut off and then again started, it may be the other tube which takes the load; or first one tube and then the other will take the load as the current drain is varied. This condition, as well as other less obvious phenomena such as a tendency for the first filter capacitor to break down regardless of its voltage rating or for rectifier tubes to have short life, results from resonance in the filter system following the high -voltage rectifier. The condition of resonance is seldom encountered in low -voltage power supplies since the capacitors used are usually high enough so that resonance does not occur. But in high voltage power supplies, where both choke inductance and filter capacitance are more expensive, the condition of resonance happens frequently. The product of inductance and capacitance which resonates at 120 cycles is 1.77. Thus a 1 -pfd. capacitor and a 1.77 henry choke will resonate at 120 cycles. In almost any normal case the LC product of any section in the filter system will be somewhat greater than 1.77, so that resonance at 120 cycles will seldom take place. But the LC product for resonance at 60 cycles is about 7.1. This is a value frequently encountered in the input section of a high- voltage power supply. It occurs with a 2 -pfd. capacitor and a choke which has 3.55 henrys of inductance at some current value. With a 2-pfd. filter capacitor following this choke, resonance will occur at the current value which causes the inductance of the choke to be 3.55 henrys. When this resonance does occur, one rectifier tube (assuming mercuryResonance www.americanradiohistory.com HANDBOOK Rectification Circuits 689 vapor types) will dim and the other will become much brighter. Thus we see that we must avoid the LC products of 1.77 and 7.1. With a swinging -type input choke, whose inductance varies over a 5 -to -1 range, we see that it is possible for resonance to occur at 60 cycles at a low value of current drain, and then for resonance to occur at 120 cycles at approximately full load on the power supply. Since the LC product must certainly be greater than 1.77 for satisfactory filtering along with peak- current limitation on the rectifier tubes, we see that with a swinging -type input choke the LC product must still be greater than 7.1 at maximum current drain from the power supply. To allow a reasonable factor of safety, it will be well to keep the LC product at maximum current drain above the value 10. It is possible to place the filter choke in the B -minus lead of the power supply which reduces the voltage potential appearing from choke winding to ground. However, the back e.m.f. of a good choke is quite high and can develop a dangerous potential from the center tap to ground on the secondary winding of the plate transformer. If the transformer is not designed to withstand this potential, it is possible to break down the insulation at this point. 31 -2 Rectification Circuits There are a large number of rectifier circuits that may be used in the power supplies for station equipment. But the simpler circuits are more satisfactory for the power levels up to the maximum permitted the radio amateur. Figure 6 shows the three most common circuits used in power supplies for amateur equipment. Half -Wave A half -wave rectifier, as shown in Rectifiers figure 6A, passes one half of the wave of each cycle of the alternating current and blocks the other half. The output current is of a pulsating nature, which can be smoothed into pure, direct current by means of filter circuits. Half -wave rectifiers produce a pulsating current which has zero output during one -half of each a -c cycle; this makes it difficult to filter the output properly into d.c. and also to secure good voltage regulation for varying loads. Full -Wove Rectifiers A full-wave rectifier consists of a pair of half -wave rectifiers working on opposite halves of the cycle, connected in such a manner that each Figure 6 MOST COMMON RECTIFIER CIRCUITS (A) shows a half -wove rectifier circuit, (8) is the standard full -wove rectifier circuit used with a dual rectifier or two rectifier tubes, and (C) is the bridge rectifier circuit. half of the rectified a -c wave is combined in the output as shown in figure 7. This pulsating unidirectional current can be filtered to any desired degree, depending upon the particular application for which the power supply is designed. A full -wave rectifier may consist of two plates and a filament, either in a single glass or metal envelope for low- voltage rectification or in the form of two separate tubes, each having a single plate and filament for high -voltage rectification. The plates are connected across the high -voltage a -c power transformer winding, as shown in figure 6B. The power transformer is for the purpose of transforming the 110 -volt a -c line supply to the desired second- www.americanradiohistory.com 690 T H Power Supplies TRANSFORMER SECONDARY VOLTAGE RECTIFIED VOLTAGE PLATE N1 RECTIFIED VOLTAGE PLATE 2 N COMBINED RECTIFIED VOLTAGE PLATES W1 0 l2 VOLTAGE + AFTER FIRST SECTION OF FILTER oL D.C. VOLTAGE AVAILABLE FOR RADIO USE Figure 7 FULL -WAVE RECTIFICATION transformer secondary voltage, the rectified output of each tube, the combined output of the rectifier, the smoothed voltage after one section of filter, and the substantially pure d.c. output of the rectifier- filter after additional sections of filter. Showing ary a -c voltages for filament and plate supplies. The transformer delivers alternating current to the two plates of the rectifier tube; one of these plates is positive at any instant during which the other is negative. The center point of the high- voltage transformer winding is usually grounded and is, therefore, at zero voltage, thereby constituting the negative B connection. While one plate of the rectifier tube is conducting, the other is inoperative, and vice versa. The output voltages from the rectifier tubes are connected together through the common rectifier filament circuit. Thus the plates alternately supply pulsating current to the output (load) circuit. The rectifier tube filaments or cathodes are always positive in polarity with respect to the plate transformer in this type of circuit. The output current pulsates 120 times per second for a full -wave rectifier connected to a 60 -cycle a -c line supply; hence the output of the rectifier must pass through a filter to smooth the pulsations into direct current. Filters are designed to select or reject alternating currents; those most commonly used in a -c power supplies are of the low -past type. Bridge The bridge rectifier (figure 6C) Rectification is a type of full -wave circuit in which four rectifier elements E R A D I O or tubes are operated from a single high -voltage winding on the power transformer. While twice as much output voltage can be obtained from a bridge rectifier as from a center- tapped circuit, the permissible output current is only one-half as great for a given power transformer. In the bridge circuit, four rectifiers and three filament heating transformer windings are needed, as against two rectifiers and one filament winding in the center- tapped full -wave circuit. In a bridge rectifier circuit, the inverse peak voltage impressed on any one rectifier tube is halved, which means that tubes of lower peak inverse voltage rating may be used for a given voltage output. Note that the center of the high voltage winding of the bridge transformer (figure 6C) is not at ground potential. Many transformers having a center -tapped winding are not designed for bridge service as the insulation between the center tap point and ground is inadequate. Lack of insulation at this point does no harm in a full -wave circuit, but may cause breakdown when the transformer is used in bridge configuration. 31 -3 Standard Power Supply Circuits Choke input is shown for all three of the standard circuits of figure 6, since choke input gives the best utilization of rectifier -tube and power transformer capability, and in addition gives much better regulation. Where greater output voltage is a requirement, where the load is relatively constant so that regulation is not of great significance, and where the rectifier tubes will be operated well within their peakcurrent ratings, the capacitor -input type of filter may be used. The capacitor -input filter gives a no -load output voltage equal approximately to the peak voltage being applied to the rectifier tubes. At full -load, the d -c output voltage is usually slightly above one -half the secondary a -c voltage of the transformer, with the normal values of capacitance at the input to the filter. With large values of input capacitance, the output voltage will run somewhat higher than the r -m -s secondary voltage applied to the tubes, but the peak current flowing through the rectifier tubes will be many times as great as the d -c output current of the power supply. The half -wave rectifier of figure 6A is commonly used with capacitor input and resistance- capacitance filter as a high -voltage supply for a cathode -ray tube. In this case the current drain www.americanradiohistory.com HANDBOOK OA Standard Circuits 691 HALF AND FULL VOLTAGE BRIDGE POWER SUPPLY BO TWO VOLTAGE BRIDGE POWER SUPPLY T osE_ +EOo,-{3- Eoo., © TWO TRANSFORMER POWER SUPPLY OD CENTER TAPPED METHOD FOR UNTAPPED TRANSFORMERS + Eoo,EL +Eoo, © TWO VOLTAGE POWER SUPPLY A FO SPECIAL FILTER CIRCUIT FOR BRIDGE RECTIFIER Figure 8 SPECIAL SINGLE PHASE RECTIFICATION CIRCUITS description of the application and operation of each of these special circuits accompanying text. is very small so that the peak -current rating of the rectifier tube seldom will be exceeded. The circuit of figure 6B is most commonly used in medium -voltage power supplies since this circuit is the most economical of filament transformers, rectifier tubes and sockets, and space. But the circuit of figure 6C, commonly called the bridge rectifier, gives better transformer utilization so that the circuit is most commonly used in higher powered supplies. The circuit has the advantage that the entire secondary of the transformer is in use at all times, instead of each side being used alternately as in the case of the full -wave rectifier. As a point of interest, the current flow through the secondary of the plate transformer is a substantially pure a -c wave as a result of better transformer utilization, instead of the pulsating d-c wave through each half of the power transformer secondary in the case of the full wave rectifier. The circuit of figure 6C will give the greatest value of output power for a given transformer weight and cost in a single -phase power supply as illustrated. But in attempting to bridge-rectify the whole secondary of a trans- is given in the former designed for a full -wave rectifier, in order to obtain doubled output voltage, make sure that the insulation rating of the transformer to be used is adequate. In the bridge rectifier circuit the center of the high -voltage winding is at a d -c potential of one -half the total voltage output from the rectifier. In a normal full -wave rectifier the center of the high -voltage winding is grounded. So in the bridge rectifier the entire high -voltage secondary of the transformer is subjected to twice the peak- voltage stress that would exist if the same transformer were used in a full -wave rectifier. High -quality full -wave transformers will withstand bridge operation quite satisfactorily so long as the total output voltage from the supply is less than perhaps 4500 volts. But inexpensive transformers, whose insulation is just sufficient for full -wave operation, will break down when bridge rectification of the entire secondary is attempted. Special Single Phase Rectification Circuits www.americanradiohistory.com Figure 8 shows six circuits which may prove valuable when it is desired to obtain more than 692 T H Power Supplies E R A D I O Eo PRIMARY Eo = 1.17 Es Is = 0.577 ID C. RIPPLE FREQUENCY= 3F RIPPLE PERCENT = 18.3 PEAK INVERSE 2.09 Eo TUBE VOLTAGE = = 2.44 Es OA 3 -PHASE STAR Figure 9 COMMON Eo. Is = POLYPHASE - 1.35 Es 0.4os ID . ' C. RIPPLE FREQUENCY = 6F RIPPLE PERCENT 4.2 PEAK INVERSE 2.09 Eo TUBE VOLTAGE 2.53 Es © 6 -PHASE STAR +Eo Eo PRIMARY Is = RECTIFICATION CIRCUITS These circuits are used when polyphase power is available for the plate supply of a high -power transmitter. The circuit at (B) is also called a full -wave three -phase rectification system. The circuits are described in the accompanying text. 2.34 Es o.81e I D.0 RIPPLE FREQUENCY =6F RIPPLE PERCENT* 4.2 PEAK INVERSE 1.03 Eo TUBE VOLTAGE . 2.44 Es © 6 -PHASE BRIDGE one output voltage from one plate transformer or where some special combination of voltages is required. Figure 8A shows a more or less common method for obtaining full voltage and half voltage from a bridge rectification circuit. With this type of circuit separate input chokes and filter systems are used on both output voltages. If a transformer designed for use with a full -wave rectifier is used in this circuit, the current drain from the full -voltage tap is doubled and added to the drain from the half-voltage tap to determine whether the rating of the transformer is being exceeded. Thus if the transformer is rated at 1250 volts at 500 ma. it will be permissible to pull 250 ma. at 2500 volts with no drain from the 1250 -volt tap, or the drain from the 1250 -volt tap may be 200 ma. if the drain from the 2500 -volt tap is 150 ma., and so forth. Figure 813 shows a system which may be convenient for obtaining two voltages which are not in a ratio of 2 to 1 from a bridge -type rectifier; a transformer with taps along the winding is required for the circuit however. With the circuit arrangement shown the voltage from the tap will be greater than one -half the voltage at the top. If the circuit is changed so that the plates of the two rectifier tubes are connected to the outside of the winding instead of to the taps, and the cathodes of the other pair are connected to the taps instead of to the outside, the total voltage ouput of the rectifier will be the same, but the voltage at the tap position will be lest than half the top voltage. An interesting variable- voltage circuit is shown in figure 8C. The arrangement may be used to increase or decrease the output voltage of a conventional power supply, as represented by transformer T,, by adding another filament transformer to isolate the filament circuits of the two rectifier tubes and adding another plate transformer between the filaments of the two tubes. The voltage contribution of the added transformer T may be subtracted from or added to the voltage produced by T1 www.americanradiohistory.com HANDBOOK Standard Circuits simply by reversing the double -pole double throw switch S. A serious disadvantage of this circuit is the fact that the entire secondary winding of transformer T_ must be insulated for the total output voltage of the power supply. An arrangement for operating a full -wave rectifier from a plate transformer not equipped with a center tap is shown in figure 8D. The two chokes L, must have high inductance ratings at the operating current of the plate supply to hold down the a -c current load on the secondary of the transformer since the total peak voltage output of the plate transformer is impressed across the chokes alternately. However, the chokes need only have half the current rating of the filter choke L2 for a certain current drain from the power supply since only half the current passes through each choke. Also, the two chokes L, act as input chokes so that an additional swinging choke is not required for such a power supply. A conventional two- voltage power supply with grounded transformer center tap is shown in figure 8E. The output voltages from this circuit are separate and not additive as in the circuit of figure 8B. Figure 8F is of advantage when it is desired to operate Class B modulators from the half- voltage output of a bridge power supply and the final amplifier from the full voltage output. Both L, and L2 should be swinging chokes but the total drain from the power supply passes through L, while only the drain of the final amplifier passes through 1.2. Capacitors C, and C2 need be rated only half the maximum output voltage of the power supply, plus the usual safety factor. This arrangement is also of advantage in holding down the "key -up" voltage of a c -w transmitter since both L, and L2 are in series, and their inductances are additive, insofar as the "critical inductance" of a choke -input filter is concerned. If 4 µfd. capacitors are used at both C, and C2 adequate filter will be obtained on both plate supplies for hum -free radiophone operation. Polyphase It Rectification Circuits cial practice in commerinstallations when the power drain from a plate supply is to be greater than about one kilowatt to use a polyphase rectification system. Such power supplies offer better transformer utilization, less ripple output and better power factor in the load placed upon the a -c line. However, such systems require a source of three -phase (or two -phase with Scott connection) energy. Several of the is usual equipment 693 more common polyphase rectification circuits with their significant characteristics are shown in figure 9. The increase in ripple frequency and decrease in percentage of ripple is apparent from the figures given in figure 9. The circuit of figure 9C gives the best transformer utilization as does the bridge circuit in the single -phase connection. The circuit has the further advantage that there is no average d -c flow in the transformer, so that three single -phase transformers may be used. A tap at half -voltage may be taken at the junction of the star transformers, but there will be d -c flow in the transformer secondaries with the power supply center tap in use. The circuit of figure 9A has the disadvantage that there is an average d -c flow in each of the windings. Rectifiers Rectifying elements in high -voltage plate supplies are almost invariably electron tubes of either the high -vacuum or mercury-vapor type, although selenium or silicon rectifier stacks containing a large number of elements are often used. Low -voltage high -current supplies may use argon gas rectifiers (Tungar tubes), selenium rectifiers, or other types of dry -disc rectification elements. The xenon rectifier tubes offer some advantage over mercury -vapor rectifiers for high- voltage applications where extreme temperature ranges are likely to be encountered. However, such rectifiers (3B25 for example) are considerably more expensive than their mercury -vapor counterparts. Peak Inverse Plate In an a -c circuit, the maxi Voltage and Peak mum peak voltage or cur Plate Current rent is V 2 or 1.41 times that indicated by the a -c meters in the circuit. The meters read the root mean- square (r.m.s.) values, which are the peak values divided by 1.41 for a sine wave. If a potential of 1,000 r.m.s. volts is obtained from a high -voltage secondary winding of a transformer, there will be 1,410 -volts peak potential from the rectifier plate to ground. In a single -phase supply the rectifier tube has this voltage impressed on it, either positively when the current flows or "inverse" when the current is blocked on the other half -cycle. The inverse peak voltage which the tube will stand safely is used as a rating for rectifier tubes. At higher voltages the tube is liable to arc back, thereby destroying or damaging it. The relations between peak inverse voltage, total transformer voltage and filter output voltage depend upon the characteristics of the filter and rectifier circuits (whether full- or half wave, bridge, single -phase or polyphase, etc.). www.americanradiohistory.com 694 T H Power Supplies =LINE VOLTS - HEATER VOLTS HEATER AMPERES + Q LINE RECTIFIER C,.., 1- CzT SELENIUM LINE RECTIFIER 1- o VOLTAGE DOUBLER FULL -WAVE e VOLTAGE DOUBLER HALF -WAVE SELENIUM RECTIFIER VOLTAGE QUADRUPLER Figure 10 TRANSFORMERLESS POWER-SUPPLY CIRCUITS Circuits such as shown above are also frequently called line -rectifier circuits. Selenium rectifiers, vacuum diodes, or gas diodes may be used as the rectifying elements in these circuits. Rectifier tubes are also rated in terms of peak plate current. The actual direct load current which can be drawn from a given rectifier tube or tubes depends upon the type of filter circuit. A full -wave rectifier with capacitor input passes a peak current several times the direct load current. In a filter with choke input, the peak current is not much greater than the load current if the inductance of the choke is fairly high (assuming full-wave rectification). A full -wave rectifier with two rectifier elements requires a transformer which delivers twice as much a -c voltage as would be the case with a half -wave rectifier or bridge rectifier. E R A D I O Mercury -Vapor Rectifier Tubes The inexpensive mercury -vapor type of rectifier tube is almost universally used in the high- voltage plate supplies of amateur and commercial transmitters. Most amateurs are quite familiar with the use of these tubes but it should be pointed out that when new or long -unused mercury -vapor tubes are first placed in service, the filaments should be operated at normal temperature for approximately twenty minutes before plate voltage is applied, in order to remove all traces of mercury from the cathode and to clear any mercury deposits from the top of the envelope. After this preliminary warm -up with a new tube, plate voltage may be applied within 20 to 30 seconds after the time the filaments are turned on, each time the power supply is used. If plate voltage should be applied before the filament is brought to full temperature, active material may be knocked from the oxide-coated filament and the life of the tube will be greatly shortened. Small r -f chokes must sometimes be connected in series with the plate leads of mercury -vapor rectifier tubes in order to prevent the generation of radio- frequency hash. These r -f chokes must be wound with sufficiently heavy wire to carry the load current and must have enough inductance to attenuate the r -f parasitic noise current to prevent it from flowing in the filter supply leads and then being radiated into nearby receivers. Manufactured mercury -vapor rectifier hash chokes are available in various current ratings from the James Millen Company in Malden, Mass., and from the J. W. Miller Company in Los Angeles. When mercury -vapor rectifier tubes are operated in parallel in a power supply, small resistors or small iron -core choke coils should be connected in series with the plate lead of each tube. These resistors or inductors tend to create an equal division of plate current between parallel tubes and prevent one tube from carrying the major portion of the current. When high vacuum rectifiers are operated in parallel, these chokes or resistors are not required. Transformerless Power Supplies Figure 10 shows a group of five different types of transformerless power supplies which are operated directly from the a -c line. Circuits of the general type are normally found in a.c. -d.c. receivers but may be used in low powered exciters and in test instruments. When circuits such as shown in (A) and (B) are operated directly from the a -c line, the rec. www.americanradiohistory.com H A N D B O O K tifier element simply rectifies the line voltage and delivers the alternate half cycles of energy to the filter network. With the normal type of rectifier tube, load currents up to approximately 75 ma. may be employed. The d -c voltage output of the filter will be slightly less than the r -m -s line voltage, depending upon the particular type of rectifier tube employed. With the introduction of the miniature selenium rectifier, the transformerless power supply has become a very convenient source of moderate voltage at currents up to perhaps 500 ma. A number of advantages are offered by the selenium rectifier as compared to the vacuum tube rectifier. Outstanding among these are the factors that the selenium rectifier operates instantly, and that it requires no heater power in order to obtain emission. The amount of heat developed by the selenium rectifier is very much less than that produced by an equivalent vacuum -tube type of rectifier. In the circuits of figure 10 (A), (B) and (C) , capacitors G and G should be rated at approximately 150 volts and for a normal degree of filtering and capacitance, should be between 15 to 60 ,dd. In the circuit of figure 10D, capacitor C, should be rated at 150 volts and capacitor G should be rated at 300 volts. In the circuit of figure 10E, capacitors C, and G should be rated at 150 volts and G and G should be rated at 300 volts. The d -c output voltage of the line rectifier may be stabilized by means of a VR tube. However, due to the unusually low internal resistance of the selenium rectifier, transform erless power supplies using this type of rectifying element can normally be expected to give very good regulation. Voltage -Doubler Figures IOC and 1OD illusCircuits trate two simple voltage- doubler circuits which will deliver a d -c output voltage equal approximately to twice the r -m -s value of the power line voltage. The no -load d -c output voltage is equal to 2.82 times the r -m -s line voltage value. At high current levels, the output voltage will be slightly under twice the line voltage. The circuit of figure IOC is of advantage when the lowest level of ripple is required from the power supply, since its ripple frequency is equal to twice the line frequency. The circuit of figure 10D is of advantage when it is desired to use the grounded side of the a -c line in a permanent installation as the return circuit for the power supply. However, with the circuit of figure IOD the ripple frequency is the same as the a -c line frequency. Standard Circuits 695 OUTSIDE COLLECTOR INSIDE COLLECTOR PHENOLIC WASNEF BASE PLATE SELENIUM COAT 100 90 - SELENIUM RECTIFIER CELL 60 Lß > 70 60 U 50 W 40 U 30 20 W 00 C, 50 100 150 200 250 300 RELATIVE LOAD CURRENT, PERCENT' OF FULL LOAD Figure 11 THE SELENIUM RECTIFIER A -The selenium rectifier is a semi -conductor stack built up of nickel plated aluminum discs coated on one side with selenium Rectifier efficiency 8- alloy. high, reaching 70"; for single phase service, dropping slightly is at high current densities. Voltage The circuit of figure 10E illustrates a voltage quadrupler circuit for miniature selenium rectifiers. In effect this circuit is equivalent to two voltage doublers of the type shown in figure 10D with their outputs connected in series. The circuit delivers a d -c output voltage under light load approximately equal to four times the r -m -s value of the line voltage. The noload d -c output voltage delivered by the quadrupler is equal to 5.66 times the r -m -s line voltage value and the output voltage decreases rather rapidly as the load current is increased. In each of the circuits in figure 10 where selenium rectifiers have been shown, conventional high -vacuum rectifiers may be substituted with their filaments connected in series and an appropriate value of the line resistor added in series with the filament string. Quadrupler 31 -4 Selenium and Silicon Rectifiers Selenium rectifiers are characterized by long life, dependability, and maintenance -free operation under severe operating conditions. The THE RADIO Power Supplies 696 POSITIVE TERMINAL NEGATIVE TERMINAL CONTACT q,yp\ww My%/ Irrror 11711. \\dA \J SILICON CELL 50 O ¶00 ¶50 LOAD CURRENT, PEPCEN7 250 200 OF FULL LOAD 300 Figure 12 VOLTAGE REGULATION OF SELENIUM CELL This graph applies to single phase lull wove bridge, and center -tap circuits which utilize both halves of the input wave. In single phase hall wave circuits the regulation will be poorer. selenium rectifier consists of a nickel -plated aluminum base plate coated with selenium over which a low temperature alloy is sprayed. The base plate serves as the negative electrode and the alloy as the positive, with current flowing readily from the base plate to the alloy but encountering high resistance in the opposite direction (figure 11A). This action results in effective rectification of an alternating input voltage and current with the efficiency of conversion dependent to some extent upon the ratio of the resistance in the conducting direction to that of the blocking direction. In normal power applications a ratio of 100 to 1 is satisfactory; however, special applications such as magnetic amplifiers often require ratios in the order of 1000 to 1. The basic selenium rectifier cell is actually a diode capable of half wave rectification. Since many applications require full wave rectification for maximum efficiency and minimum ripple, a plurality of cells in series, parallel, or series -parallel combinations are stacked in an assembly. Selenium rectifiers are operated over a wide range of voltages and currents. Typical applications range from a few volts at milliamperes of current to thousands of amperes at relatively high voltages. The efficiency of high quality selenium rectifiers is high, usually in the order of 90% in three phase bridge circuits and 70% in single phase bridge circuits. Of particular interest is the very slight decrease in efficiency even at high current overloads (figure 11B). Threshold Voltage and Aging A minimum voltage is required to permit a selen- ium rectifier to conduct in the forward direction. This voltage, commonly known as the threshold voltage, precludes the use of selenium rectifiers at ex- SPRING Figure 13 THE SILICON CELL The common silicon rectifier is a pressure contact device capable of operation in ambient temperatures as high as 150 °C. Heavy end ferrules that fit standard fuse clips are large enough to provide "heat sink" action. The positive ferrule is grooved to provide polarity identification and prevent incorrect mounting. tremely low ( less than one volt) applications. The threshold voltage will vary with temperature and will increase with a decrease in temperature. Under operating conditions, and to a lesser extent when idle, the selenium rectifier will age. During the aging period the forward resistance will gradually increase, stabilizing at a new, higher value after about one year. This aging will result in approximately a 7% decrease in output voltage. Voltage Regulation The selenium rectifier has extremely low internal impedance which exhibits non -linear characteristics with respect to applied voltage. This results in good voltage regulation even at large overload currents. Figure 12 shows that as the load is varied from zero to 300% of normal, the output voltage will change about 10 %. It should be noted that because of non -linear characteristics, the voltage drop increases rapidly below 50% of normal load. Of all recent developments in the field of semi -conductors, silicon rectifiers offer the most promising range of applications; from extreme cold to high temperature, and from a few watts of output power to very high voltage and currents. Inherent characteristics of silicon allow junction temperatures in the order of 200 °C before the material exhibits intrinsic properties. This extends the operating range of silicon devices beyond that of any other efficient semi -conductor and the excellent thermal range coupled with very small size per watt of output power make silicon rectifiers applicable where other rectifiers were previously considered impractical. Silicon Rectifiers Silicon Current Density www.americanradiohistory.com The current density of a silicon rectifier is very high, and on present designs ranges HANDBOOK Mobile Power Supply 697 from 600 to 900 amperes per square inch of effective barrier layer. The usable current density depends upon the general construction of the unit and the ability of the heat sink to conduct heat from the crystal. The small size of the crystal is illustrated by the fact that a rectifier rated at 15 amperes d.c., and 150 amperes peak surge current has a total cell volume of only .00023 inches. Peak currents are extremely critical because the small mass of the cell will heat instantaneously and could reach failure temperatures within a time lapse of microseconds. The assembly of a typical silicon cell is shown in figure 13. The reverse direction of a silicon rectifier is characterized by extremely high resistance, up to 10" ohms below a critical voltage point. This point of avalanche voltage is the region of a sharp break in the resistance curve, followed by rapidly decreasing resistance (figure 15A). In practice, the peak inverse working voltage is usually set at least 20% below the avalanche point to provide a safety factor. The forward direction, or direction of low resistance determines the majority of power loss within the semi -conductor device. Figure 15B shows the static forward current characteristics versus applied voltage. The threshold voltage is about 0.6 volts. Since the forward resistance of a semi-conductor is very low, any unbalance between threshold voltages or internal voltage drop would cause serious unbalance of load distribution and ultimate failure of the overloaded section. A small resistance should therefore be placed in series with each half wave section Operating Characteristics Figure 14 MINIATURE SEMI- CONDUCTOR TYPE RECTIFIER Raytheon CK -777 power rectifier bolts to chassis to gain large "heat sink" area. Low internal voltage drop and high efficiency permit small size of unit. operating in parallel to balance the load currents. Some interesting and practical semi- conductor power supplies are shown in figure 16. Remember that the circuits of figure 16A and B, and those of figure 10 are "hot" with respect to one side of the power line. 31 -5 100 Watt Mobile Power Supply High efficiency and compact size are the most important factors in the design of mobile power supplies. The power package described in this section meets these stringent require- W o: W t 1 00 W K ! Vt O4 to WF >z FW .. HANDBOOK Decibel -Power Conversion ...[.z.Z,...--M....-=MN 60 .r.RWz - Rtt.t...w!T es1.- 50 -_ - 40 y_y_ .._.....a.t,.as,- 40 30 20 IO ....trL _,,..... I..t.t... .....liaa, ...... -tl.t....lT.r:in/a! --.i tift....R.....lace .... R.t....riarr= ._._._._..._ ....... =INN m.. t.fiti..tIR11Llbkr.mmilmm. - - EE.- T...... ...... - _..-----..s__.. _..-it.ii it... .--- -t. = - =..w!}T71.IIIg1ifJiC m.m..RL>ai'0.1116 Num_u.: -a-am... .i...}7TiI,1------80,....---. .t .t.. , . -50 -60 ..ri,ulsu/.a- -70 -80 o MI M1111111111110 = i=M.iiTT.TÜ,INIGi'yf. ti=t...Rrlsa - 1. - ti 19 -70 MI MOW t.t...ttl-twncr.trn t.titit...irdt.la:Jl:. =:>teiAarli -_ aims . 9 1 2 POWER r are ht - IN I11 _GEE -80 MO 3 4 -90 6 Bard an .eei reW Figure 9. CONVERSION CHART: POWER TO DECIBELS Power levels between 6 micromicrowatts and 6000 watts may be referred to corresponding decibel levels between -90 and 60 db, and vice versa, by means of the above chart. Fifteen ranges are provided. Each curve begins at the same point where the preceding one ends, enabling uninterrupted go of the wide db and power ranges with condensed chart. For example: the lowermost curve ends at -00 db or 60 micromlcrowatts and the next range starts at the same level. Zeno db level is taken as 6 milliwatts (.006 watt). www.americanradiohistory.com J W Il sltaer: I.t... -20 MI INIMMIMI..ll. Rtta.. t.MM..11i\+It61= itit.t...m _-- .e.r.riTr:1SIl Rtt.t...IR iLLrr..MMt.IIIN..e .....R1.1tii ee17üI IO _.r I -1-40 20 = ll'S' MMI. Ms 19.=.....mol. 30 . .rTS1/Jii t.elt.t...i ----..R1P 773 It is often convenient to he able to convert a decibel value to a power equivalent. The formula used for this operConverting Decibels to Power ation THE RADIO Radio Mathematics and Calculations 774 Solution: --17.3 2.7 - 20 is P = Nan 10 0.006 X antilog N'ö where P is the desired level in watts and Nab the decibels to be converted. To determine the power level P from a decibel equivalent, simply divide the decibel value by 10; then take the number comprising the antilog and multiply it by 0.006; the product gives the level in watts. Note: In problems dealing with the conversion of miner decibels to power, it often happens that the decibel value -Na, is not divisible by 10. When this is the case, the numerator in the factor - - N,,n 10 -- must be made evenly divisible by 10, the negative signs must be observed, and the quotient labeled accordingly. To make the numerator evenly divisible by 10 proceed as follows: Assume, for example, is some such value as -38; to that make this figure evenly divisible by 10, we must add -2 to it, and, since we have added a negative 2 to it, we must also add a positive 2 so as to keep the net result the same. Our decibel value now stands, -40 + 2. Dividing both of these figures by 10, as in the 4 and +0.2. Putequation above, we have ting the two together we have the logarithm -4.2 with the negative characteristic and the positive mantissa as required. The following examples will show the technique to be followed in practical problems. (a) The output of a certain device is rated at -74 db. What is the power equivalent? Solution: - N,ib - -74 not evenly divisible NÖ, ( Routine: -- 80 74 +6 6 Nni 10 - by 10) +6 -80 + 10 6 - 8.6 ontilog -8.6 = 0.000 000 04 .006 X 0.000 000 04 = 0.000 000 000 24 watt or 240 micro- microwatt (b) This example differs somewhat from that of the foregoing one in that the mantissas are added differently. A low- powered amplifier has an input signal level of -17.3 db. How many milliwatts does this value represent? - -20 + Antilog + 2.7 2.7 + 2.7 -2.27 10 -2.27 = 0.0186 0.006 X 0.0186 = 0.000 0.1116 milliwatt 1 1 16 watt or Input voltages: To determine the required input voltage, take the peak voltage necessary to drive the last class A amplifier tube to maximum output, and divide this figure by the total overall voltage gain of the preceding stages. Computing Specifications: From the preceding explanations the following data can be computed with any degree of accuracy warranted by the circumstances: (1) Voltage amplification (2) Overall gain in db (3) Output signal level in db (4) Input signal level in db (5) Input signal level in watts (6) Input signal voltage When a power level is available which must be brought up to a new power level, the gain required in the intervening amplifier is equal to the difference between the two levels in decibels. If the required input of an amplifier for full output is -30 decibels and the output from a device to be used is but -45 decibels, the pre -amplifier required should have a gain of the difference, or 15 decibels. Again this is true only if the two amplifiers are properly matched and no losses are introduced due to mismatching. Push -Pull Amplifiers To double the output of any cas cade amplifier, it is only necessary to connect in push -pull the last amplifying stage, and replace the inter stage and output transformers with push -pull types. To determine the voltage gain (voltage ratio) of a push -pull amplifier, take the ratio of one hall of the secondary winding of the push. pull transformer and multiply it by the of one of the output tubes in the push -pull stage; the product, when doubled, will be the voltage amplification, or step-up. a Other Units and Zero Levels When working with deci- bels one should not immediately take for granted that the zero level is 6 milliwatts for there are other zero levels in use. In broadcast stations an entirely new system is now employed. Measurements made in www.americanradiohistory.com HANDBOOK Trigonometry acoustics are now made with the standard zero level of 10-'° watts per square cm. Microphones are often rated with reference to the following zero level: one volt at open circuit u hen the sound pressure is one millibar. In any case, the rating of the microphone must include the loudness of the sound. It is obvious that this zero level does not lend itself readily for the calculation of required gain in an am- plifier. The VU: So far, the decibel has always referred to a type of signal which can readily be measured, that is, a steady signal of a single frequency. But what would be the power level of a signal which is constantly varying in volume and frequency? The measurement of voltage would depend on the type of instrument employed, whether it is measured with a thermal square law meter or one that shows average values; also, the inertia of the movement will change its indications at the peaks and valleys. After considerable consultation, the broadcast chains and the Bell System have agreed on the VU. The level in VU is the level in milltu att zero level and measdecibels above ured with a carefully defined type of instrument across a 600 ohm line. So long as we deal with an unvarying sound, the level in VU milliwatt; but is equal to decibels above when the sound level varies, the unit is the VU and the special meter must be used. There is then no equivalent in decibels. The Neper: We might have used the natural logarithm instead of the common logarithm when defining our logarithmic unit of sound. This was done in Europe and the unit obtained is known as the neper or napier. It is still found in some American literature on filters. 1 1 1 1 neper = 8.686 decibels decibel = 0.1151 neper AC Meters With Decibel Scales test instruments are now equipped with scales calibrated in deci- Many bels which is very handy when making measurements of frequency characteristics and gain. These meters are generally calibrated for connection across a 500 ohm line and for a zero level of 6 milliwatts. When they are connected across another impedance, the reading on the meter is no longer correct for the zero level of 6 milliwatts. A correction factor should be applied consisting in the addition or subtraction of a steady figure to all readings on the meter. This figure is given by the equation: db to be added = 10 log szo where Z is the impedance of the circuit under measurement. SECOND FIRST QUADRANT QUADRANT THIRD FOURTH QUADRANT QUADRANT 775 Figure 10. CIRCLE IS DIVIDED INTO THE FOUR QUADRANTS BY TWO PER- PENDICULAR LINES AT RIGHT ANGLES TO EACH OTHER. "northeast" quadrant thus formed is known os the first quadrant; the others are The numbered consecutively in a counterclockwise direction. Trigonometry Trigonometry is the science of mensuration of triangles. At first and Use glance triangles may seem to have little to do with electrical phenomena; however, in a.c. work most currents and voltages follow laws equivalent to those of the various trigonometric relations which we are about to examine briefly. Examples of their application to a.c. work will he given in the Definition section on Vectors. Angles are measured in degiees or in radians. The circle has been divided into 360 degrees, each degree into 60 minutes, and each minute into 60 seconds. A decimal division of the degree is also in use because it makes calculation easier. Degrees, minutes and seconds are indicated by the following signs: °, ' and " Example: 6° 5' 23" means six degrees, five minutes, twenty -three seconds. In the decimal notation we simply write 8.47 °, eight and forty -seven hundredths of a degree. When a circle is divided into four quadrants by two perpendicular lines passing through the center (Figure 10) the angle made by the two lines is 90 degrees, known as a right angle. Two right angles, or 180° equals a straight angle. The radian: If we take the radius of a circle and bend it so it can cover a part of the circumference, the arc it covers subtends an angle called a radian (Figure 11 ). Since the diameter, of a circle equals 2 times the radius, there are 27 radians in 360 °. So we have the following relations: 7= 3.14159 1 radian =57° 17'45 " = 57.2958° 1 degree = 0.01745 radians /2 radians =90° yr radians =180 °' 7/3 radians =60° www.americanradiohistory.com 776 THE RADIO Radio Mathematics and Calculations In the angle A, Figure 13A, a line is drawn from P, perpendicular to b. Regardless of the point selected for P, the ratio a/c will always be the same for any given angle, A. So will all the other proportions between a, b, and c remain constant regardless of the position of point P on c. The six possible ratios each are named and defined as follows: a sine A = Figure 11. THE RADIAN. tangent radian is an angle whose arc is exactly equal to the length of either side. Note that the angle is constant regardless of the length of the side and the arc so long as they are equal. A radian equals 57.2958 A a = A . In trigonometry we consider an angle generated by two lines, one stationary and the other rotating as if it were hinged at 0, Figure 12. Angles can be greater than 180 degrees and even greater than 360 degrees as illustrated in this figure. Two angles are complements of each other when their sumis 90 °, or a right angle. A is the complement of B and B is the complement of A when A=(90° -B) cosine A c cosecant A = sin. 60° _ a t/2 1(3 = a and a 1/2 =bc = 1/2 1/2 1/2 cot 60° - = 1 = 1/2 sec 60° = b 1 = 1/2 =2 csc 60° = a 1 = IS 2/j 1/2 Another example: Let the angle be 45 °, then the relations between the lengths of a, b, and c are as shown in Figure 13C, and the six functions are: AC AN ANGLE =1/2 1/2 V-3 tan 60° = b = Two angles are supplements of each other when their sum is equal to'a straight angle, or 180 °. A is the supplement of B and B is the supplement of A when B= (180 ° -A) = cos 60 ° -A) A= (180 ° -B) c Let us take a special angle as an example. For instance, let the angle A be 60 degrees as in Figure 13B. Then the relations between the sides are as in the figure and the six functions become: and when B= (90° b cotangent A = C secant A = = Figure 12. GENERATED BY TWO LINES, ONE STATIONARY AND THE OTHER ROTATING. stationary; the line with the small arrow at the far end rotates in a counterclockwise direction. At the position illustrated in the lefthandmost section of the drawing it makes an angle, A, which is less thon 90 and is therefore in the first quadrant. In the position shown in the second portion of the drawing the angle A has increased to such a value that it now lies in the third quadrant; note that an angle can be greater than 180 . In the third illustration the angle A is in the fourth quadrant. In the fourth position the rotating vector has made more than one complete revolution and is hence in the fifth quadrant; since the fifth quadrant is an exact repetition of the first quadrant, its values will be the same os in the lefthandmost aortion of the illustration. The line OX is www.americanradiohistory.com Trigonometric Relations H A N D B O O K 777 P C=e 90" O b=0 Figure 13. THE TRIGONOMETRIC FUNCTIONS. In the right triangle shown b and c; the trigonometric sides a, b and c. In (B) are is 1. In (C) angle A is IS ; sin 45° in (A) the side opposite the ongle A is o, while the adjoining sides are functions of the ongle A are completely defined by the ratios of the shown the lengths of the sides a and b when angle A is 60- and side c a and b equal 1, while c equals V-2. In (D) note that c equals a for a right angle while b equals O. = 1/, V-2 tan 45° = 1 = 1z Ari 1 = cot 45° = 1 = sec 45° °= --= sec a = b - a 0 cos 12 _ cos A c c co b When the angle values are then: sin 0 sec 0° a c = c b c = 1 c 0 a - o = o C _ -a = cosec 90° is zero, a -= tan 0° = b0 A -cot A =0 =sin (complement of A) =sin (90 ° -A) the right triangle of Figure 15, A =b /c =sin B and B=90° -A or the complement of A. For the same reason: because in cos cotA = tan (90 ° -A) csc A = =0 and b = 0 cos 0° 0 cot 0° = b =c identities: tan A = a/b cot A = b/a = = a b b a tan A cot A 1 CO In the same triangle we can do the same for functions of the angle B 0 a cosec0° = sin A For the same reason we have the following 1 =c. The = c -A) (90 ° sec In the right triangle of Figure 15, sin A =a /c and by transposition Relations in Right Triangles a -=-= 90° b 90° = cot 90° = = 1 sec From the definitions also follows the relation a 1 A - cos and tan A 1 There are some special difficulties when the angle is zero or 90 degrees. In Figure 13D an angle of 90 degrees is shown; drawing a line perpendicular to b from point P makes it fall on top of c. Therefore in this case a = c and b = 0. The six ratios are now: tan 90° I cosec A 1 V-2- cosec 45° sin 90 sin A = 1 = 1 - _ cos 45° It follows from the definitions that Relations Between Functions 1 _ c a _c 0- co In general, for every angle, there will be definite values of the six functions. Conversely, when any of the six functions is known, the angle is defined. Tables have been calculated giving the value of the functions for angles. From the foregoing we can make up a small table of our own (Figure 14), giving values of the functions for some common angles. Angle Sin Cos. Tan Cot Sec. 0 0 1 0 m 1 30° 45° 1/2 60° 1/2 90° 1/2 1/2 \ 13 1 1/2 0 1/3 \ VT 3 1/3 V-3- 1 1/2 1 1/3 w 0 v, \ri 2/3 2 Cosec. 2 V 2/3 cc Figure 14. Values of trigonometric functions for common angles in the first quadrant. www.americanradiohistory.com ' VT 1 THE RADIO Radio Mathematics and Calculations 778 POSITIVE FUNCTIONS QUADRANT FIRST SECOND QUADRANT sins, cosec all functions b Figure 15. in this figure the sides a, b, and c are used to define the trigonometric functions of angle B as well as angle A. sin B = b/c cos B = a/c tan B = b/a b=csin cot B a = a/b Functions of Angles Greater than 90 Degrees a b = = = B tan, cot b cot B In angles greater than 90 degrees, the values of a and b become negative on occasion in ac- THIRD QUADRANT = tan A = sec a = pos. cos -b = neg. cot A A = -c neg. c a cosec A = = A = -b - c a = tan A = b-a -a = neg. cos = pos. cot A = A = sec sin A -b = neg. = pos. = tan A = = pos. -b A = = neg. cosec A = - -c a = neg. And in the fourth quadrant (270° to 360 °): neg. For an angle in the third quadrant (180° to 270 °), the functions are sin A FUNCTIONS. The functions listed in this diagram are positive; all other functions are negative. neg. --ab c = FOURTH QUADRANT Figure 17. SIGNS OF THE TRIGONOMETRIC cordance with the rules of Cartesian coordinates. When b is measured from 0 towards the left it is considered negative and similarly, when a is measured from 0 downwards, it is negative. Referring to Figure 16, an angle in the second quadrant (between 90° and 180 °) has some of its functions negative: sin A cosine, secant c cos B a tan B sec A = -- a c -a b = neg. = neg. -b cos pos. = pos. = neg. =-a = neg. cot A = c cosec b A = A -ba Summarizing, the sign of the functions in each quadrant can be seen at a glance from Figure 17, where in each quadrant are written the names of functions which are positive; those not mentioned are negative. SECOND QUADRANT tb FOURTH THIRD QUADRANT QUADRANT Figure 16. TRIGONOMETRIC FUNCTIONS IN THE SECOND, THIRD, AND FOURTH QUADRANTS. The trigonometric functions in these quadrants are similar to first quadrant values, but signs of the functions vary as listed in the text and in Figure 17. www.americanradiohistory.com the HANDBOOK 90 A1 270' 180' Figure 18. SINE AND COSINE CURVES. In (A) we have a sine curve drawn in Cartesian coordinates. This is the usual representation of an alternating current wave without substantial harmonics. In (B) we have a cosine wave; note that it is exactly similar to a sine wave displaced by 90 or n 2 radians. 90" The sine u ate. When we have the relation y= sin x. where x is an angle measured in radians or degrees, we can draw a curve of y versus x for all values of the independent variable, and thus get a good conception how the sine varies with the magnitude of the angle. This has been done in Figure 18A. We can learn from this curve the following facts. 1. The sine varies between +1 and -1 2. It is a periodic curve, repeating itself after every multiple of 27 or 360° 3. Sin x = sin (180 ° or sin (ir 4. Sin x = -sin (180° + x), or Graphs of Trigono- 3. -x) -sin srr 360" -- (R + x) The cosine :cave. Making a curve for the function y = cos x, we obtain a curve similar to that for y = sin x except that it is displaced by 90° or 7/2 radians with respect to the Y -axis. This curve (Figure 18B) is also periodic but it does not start with zero. We read from the curve: 1. The value of the cosine never goes beyond +1 or -1 2. The curve repeats, after every multiple of 27 radians or 360° == = Cos x 7r 450" - -- - Ado - - - Now LI =_ 2 = -cos (180° -x) -x) Irr or -x) -x) r cos (360 ° or cos (tir The graph of the tangent is illustrated in Figure 19. This is a discontinuous curve and illustrates well how the tangent increases from zero to infinity when the angle increases from zero to 90 degrees. Then when the angle is further increased, the tangent starts from minus infinity going to zero in the second quadrant, and to infinity again in the third quadrant. 1. The tangent can have any value between 4. Cos x +m 2. 3. 4. and - The curve repeats and the period is err radians or 180 °, not 2,r radians Tan x = tan (180° +x) or tan (-rr +x) Tan x = -tan (180° or -tan -x) (- -x) The graph of the cotangent is the inverse of that of the tangent, see Figure 20. It leads us to the following conclusions: 1. The cotangent can have any value be- - tween + 2. It periodic curve, the period being 3. 4. radians or 180° cot (180° +x) or cot (^.r Cot x cot (180° or Cot x = is a -cot m and - -x) (r -x) 270° 720' 540° 450 630° The tangent curve increases from 0 to m with an angular increase of 90 °. In the next 180' increases from -m to + 180° 630' 450° 270' 90° 0° 360 540° 720' Figure 20. COTANGENT CURVES. Figure 19. TANGENT CURVES. it +x) en" 360° 180° C - - _ 720' 630° 540 - z.r 90' 720' 630 "37r -cos (7 metric Functions -x) S 2m - o1m 540' 450 360 270" 180" _ (B) 0° 779 Trigonometric Curves s- Cotangent curves are the inverse of the tanin gent curves. They vary from + m to each pair of quadrants. www.americanradiohistory.com -m 780 THE RADIO Rodio Mathematics and Calculations Figure 22. Vectors may be added as shown in these sketches. In each case the long vector represents the vector sum of the smaller vectors. For many engineering applications sufficient accuracy can be obtained by this method which avoids long and laborious calculations. COSINE Figure 21. ANOTHER REPRESENTATION OF TRIGONOMETRIC FUNCTIONS. If the radius of a circle is considered as the unit of measurement, then the lengths of the various lines shown in this diagram are numerically equal to the functions marked adjacent to them. The graphs of the secant and cosecant are of lesser importance and will not be shown here. They are the inverse, respectively, of the cosine and the sine, and therefore they vary from +1 to infinity and from -1 to infinity. Perhaps another useful way of visualizing the values of the functions is by considering Figure 21. If the radius of the circle is the unit of measurement then the lengths of the lines are equal to the functions marked on them. - Trigonometric Tables There are two kinds of trigonometric tables. The first type gives the functions of the angles, the second the logarithms' of the functions. The first kind is also known as the table of natural trigonometric functions. These tables give the functions of all angles between 0 and 45 °. This is all that is necessary for the function of an angle between 45° and 90° can always be written as the co- function of an angle below 45 °. Example: If we had to find the sine of 48 °, we might write sin 48° = cos (90° -48 °) = cos 42° Tables of the logarithms of trigonometric functions give the common logarithms (log.) of these functions. Since many of these logarithms have negative characteristics, one should add -10 to all logarithms in the table which have a characteristic of 6 or higher. For instance, the log sin 24° = 9.60931 -10. Log tan 1° = 8.24192 -10 but log cot 1° = 1.75808. When the characteristic shown is less than 6, it is supposed to be positive and one should not add -10. Vectors A ¡calar quantity has magnitude only; a vector quantity has both magnitude and direction. When we speak of a speed of 50 miles per hour, we are using a scalar quantity, but when we say the wind is Northeast and has a velocity of 50 miles per hour, we speak of a vector quantity. Vectors, representing forces, speeds, displacements, etc., are represented by arrows. They can be added graphically by well known methods illustrated in Figure 22. We can make the parallelogram of forces or we can simply draw a triangle. The addition of many vectors can be accomplished graphically as in the same figure. In order that we may define vectors algebraically and add, subtract, multiply, or divide them, we must have a logical notation system that lends itself to these operations. For this purpose vectors can be defined by coordinate systems. Both the Cartesian and the polar coordinates are in use. Vectors Defined by Cartesian Coordinates Since we have seen how the sum of two vectors is obtained, it follows from Figure 23, that the vector Z equals the sum of the two vectors x and y. In fact, any vector can be resolved into vectors along the X- and Y-axis. For convenience in working with these quantities we need to dis- yS YA5 o 3 Figure 23. RESOLUTION OF VECTORS. i Any vector such as may be resolved Into two vectors, x and y, along the X- and Yaxes. If vectors are to be added, their respective z and y components may be added to find the x and y components of the resultant vector. www.americanradiohistory.com HANDBOOK Vectors Addition of Vectors 781 An examination of Figwill show that ure 24 the two vectors =x,+jY, Z may adding or be added subtracting or their subtracted x or y R by com- ponents separately. tinguish between the X- and Y- component, and so it has been agreed that the Y- component alone shall be marked with the letter j. Example (Figure 23) : Z =3 +4j Note again that the sign of components along the X -axis is positive when measured from 0 to the right and negative when measured from O towards the left. Also, the component along the Y -axis is positive when measured from 0 upwards, and negative when measured from 0 downwards. So the vector, R, is described as R =5 -3j Vector quantities are usually indicated by some special typography, especially by using a point over the letter indicating the vector, as R. Absolute Value of a Vector The absolute or scalar value of vectors such as 2 or R in Figure 23 is easily found by the theorem of Pythagoras, which states that in any right -angled triangle the square of the side opposite the right angle is equal to the sum of the squares of the sides adjoining the right angle. In Figure 23, OAB is a right -angled triangle; therefore, the square of OB (or Z) is equal to the square of OA (or x) plus the square of AB (or y). Thus the absolute values of Z and R may be determined as follows: IZI= Vx' +y' ZI= V3' +4' =5 RI= b +3'= x: + = Z j Y= + x, R -Z = x, - The vertical lines indicate that the absolute or scalar value is meant without regard to sign or direction. + j + (y, Y=) x: + - (y, j y2) Let us consider the operator j. If we have a vector a along the X -axis and add a j in front of it (multiplying by j) the result is that the direction of the vector is rotated forward 90 degrees. If we do this twice (multiplying by 11) the vector is rotated forward by 180 degrees and now has the value -a. Therefore multiplying by is equivalent to multiplying by -1. Then f = -1 12 and j = V -1 This is the imaginary number discussed before under algebra. In electrical engineering the letter j is used rather than i, because i is already known as the symbol for current. Multiplying Vectors When two vectors are to be multiplied we can perform the operation just as in algebra, remembering that j2 _ -1. RZ= (x, +j',) =x,x2 4- ¡XI = - y, Y= x, x: +jy2) (x : Y:4-jx_yt+j'y,ya + j (x, Ya + x: y,) Division has to be carried out so as to remove the j -term from the denominator. This can be done by multiplying both denominator and numerator by a quantity which will eliminate j from the denominator. Example: R x, +jy,_(x, +jy,) Z x: + (x: hY: + YiY= + x:' =5.83 x: For the same reason we can carry out subtraction by subtracting the horizontal components and subtracting the vertical components x,x: 34 + can be added, if we add the X- components and the Y- components separately. Figure 24. ADDITION OR SUBTRACTION OF VECTORS. Vectors = + j (x:y, + (x: jY:) (x: -jxa) - jy:) - x,y:) Ya' A vector can also be defined in polar coordinates by its magnitude and its vectorial angle with an arbitrary reference axis. In Figure 25 Polar Coordinates www.americanradiohistory.com THE RADIO Radio Mathematics and Calculations 782 Figure 26. Vectors can be trans formed from Cartesian into polar notation as shown in this figure. X Figure 25. IN THIS FIGURE A VECTOR HAS BEEN REPRESENTED IN POLAR INSTEAD OF CARTESIAN COORDINATES. In polar coordinates a vector is defined by a magnitude and an angle, called the vectorial angle, instead of by two magnitudes as in Cartesian coordinates. the vector Z has a magnitude 50 and a vectorial angle of 60 degrees. This will then be written i= 5OL6O° A vector a + jb can be transformed into polar notation very simply (see Figure 26) =o +jb= Vo'+ b'Lton'u In this connection tan -' means the angle of which the tangent is. Sometimes the notation arc tan b/a is used. Both have the same meaning. A polar notation of a vector can be transformed into a Cartesian coordinate notation in the following manner (Figure 27) i = pLA = p cos A + jp sin A A sinusoidally alternating voltage or current is symbolically represented by a rotating vector, having a magnitude equal to the peak voltage or current and rotating with an angular velocity of 2-,rf radians per second or as many revolutions per second as there are cycles per which flows due to the alternating voltage is not necessarily in step with it. The rotating current vector may be ahead or behind the voltage vector, having a phase difference with it. For convenience we draw these vectors as if they were standing still, so that we can indicate the difference in phase or the phase angle. In Figure 28 the current lags behind the voltage by the angle e, or we might say that the voltage leads the current by the angle B. Vector diagrams show the phase relations between two or more vectors (voltages and currents) in a circuit. They may be added and subtracted as described; one may add a voltage vector to another voltage vector or a current vector to a current vector but not a current vector to a voltage vector (for the same reason that one cannot add a force to a speed). Figure 28 illustrates the relations in the simple series circuit of a coil and resistor. We know that the current passing through coil and resistor must he the same and in the same phase, so we draw this current I along the X -axis. We know also that the voltage drop IR across the resistor is in phase with the current, so the vector IR representing the voltage drop is also along the X-axis. The voltage across the coil is 90 degrees ahead of the current through it; /X must therefore be drawn along the Y -axis. E the applied voltage must be equal to the vectorial sum of the two voltage drops, IR and IX, and we have so constructed it in the drawing. Now expressing the same in algebraic notation, we have =IR+jIX second. The instantaneous voltage, e, is always equal to the sine of the vectorial angle of this rotating vector, multiplied by its magnitude. e = E IZ Dividing by I sin 2^rft The alternating voltage therefore varies with time as the sine varies with the angle. If we plot time horizontally and instantaneous voltage vertically we will get a curve like those in Figure 18. In alternating current circuits, the current = IR + jIX i =R +jX Due to the fact that a reactance rotates the voltage vector ahead or behind the current vector by 90 degrees, we must mark it with a j in vector notation. Inductive reactance will have a plus sign because it shifts the voltage vector forwards; a capacitive reactance is neg- www.americanradiohistory.com HANDBOOK p Graphical Representation COS A Figure 27. Vectors can be trans formed from polar into Cartesian notation as shown in this figure. ative because the voltage will lag behind the current. Therefore: X,. = + X =_j Graphical Representation 1 2nfC In Figure 28 the angle 8 is known as the phase angle between E and I. When calculating power, only the real components count. The power in the circuit is then = P but IR OR) I = P= E = tan 8 EIcost) 8 = Formulas and physical laws are often presented in graphical form; this gives us a "bird's eye view" of various possible conditions due to the variations of the quantities involved. In some cases graphs permit us to solve equations with greater ease than ordinary algebra. Coordinate Systems cos This coi 8 is known as the power factor of the circuit. In many circuits we strive to keep the angle e as small as possible, making cos e as near to unity as possible. In tuned circuits, we use reactances which should have as low a power factor as possible. The merit of a coil or condenser, its Q. is defined by the tangent of this phase angle: Q Figure 28. VECTOR REPRESENTATION OF A SIMPLE SERIES CIRCUIT. The righthand portion of the illustration shows the vectors representing the voltage drops in the coil and resistance illustrated at the left. Note that the voltage drop across the coil Xi. leads that across the resistance by 90 °. -fL j 2 783 X/R For an efficient coil or condenser, Q should large as possible; the phase -angle should then be as close to 90 degrees as possible, making the power factor nearly zero. Q is almost but not quite the inverse of cos e. Note that in be as All of us have used coordinate systems with- out realizing it. For instance, in modern cities we have numbered streets and numbered avenues. By this means we can define the location of any spot in the city if the nearest street crossings are named. This is nothing but an application of Cartesian coordinates. In the Cartesian coordinate system (named after Descartes), we define the location of any point in a plane by giving its distance from each of two perpendicular lines or axes. Figure 30 illustrates this idea. The vertical axis is called the Y -axis, the horizontal axis is the X -axis. The intersection of these two axes is called the origin, O. The location of a point, P, (Figure 30) is defined by measuring the respective distances, x and y along the X -axis and the Y -axis. In this example the distance along the X -axis is 2 units and along the Yaxis is 3 units. Thus we define the point as Figure 29 Q =X /R and cos e = 0= TAN e. R/Z When Q is more than 5, the power factor is than 20%; we can then safely say Q = /cos e with a maximum error of about 21/2 percent, for in the worst case, when cos 8 = 0.2, Q will equal tan e = 4.89. For higher values of Q, the error becomes less. Note that from Figure 29 can be seen the simple relation: POWER FACTOR =COS e- less 1 -R+jX, IZI= R'+Xt' Figure 29. The figure of merit of a coil and its resistance represented by the ratio of the inductive reactance to the resistance, which as shown is in this diagram is equal to R' which equals tan 9. For large values of 0 (the phase angle) this is approximately equal to the reciprocal of the cos O. - 784 Radio Mathematics and Calculations Y 2000 7 1800 Ile SECOND QUADRANT FIRST 1600 QUADRANT 5 4 a 1400 1200 1000 2 ROO 600 X8-7- 6- 5- 4 -3 -2 -I 100 I 01 2 3 4 5 6 7 3 1 TH RD QUADRANT Ill - S 4 5 FOURTH 6 QUADRANT p Y III Figure 30. CARTESIAN COORDINATES. The location of any point can be defined by its distance from the X and Y axes. 3 or we might say x = 2 and y = 3. The measurement x is called the abscissa of the point and the distance y is called its ordinate. It is arbitrarily agreed that distances measured from 0 to the right along the X -axis shall be reckoned positive and to the left negative. Distances measured along the Y -axis are positive when measured upwards from 0 and negative when measured downwards from 0. This is illustrated in Figure 30. The two axes divide the plane area into four parts called quadrants. These four quadrants are numbered as shown P 2, in the figure. It follows from the foregoing statements, that points lying within the first quadrant have both x and y positive, as is the case with the point P. A point in the second quadrant has a negative abscissa, x, and a positive ordinate, y. This is illustrated by the point Q, which has the coordinates x = -4 and y = +1. Points in the third quadrant have both x and y negative. x = -5 and y = -2 illustrates such a point, R. The point S, in the fourth quadrant has a negative ordinate, y and a positive abscissa or x. In practical applications we might draw only as much of this plane as needed to illus- trate our equation and therefore, the scales along the X -axis and Y -axis might not start with zero and may show only that part of the scale which interests us. Representation of Functions In the equation: 300,000 200 300 400 600 8 2 IR THE RADIO Figure 31. REPRESENTATION OF A SIMPLE FUNCTION IN CARTESIAN COORDINATES. 300,000 In this chart of the function fk, = mr distances along the X axis represent wavelength in meters, while those along the Y axis represent frequency in kilocycles. A curve such as this helps to find values between those calculated with sufficient accuracy for most purposes. f is said to be a function of X. For every value of f there is a definite value of X. A variable is said to be a function of another variable when for every possible value of the latter, or independent variable, there is a definite value of the first or dependent variable. For instance, if y = 5x', y is a function of x and x is called the independent variable. When a = 3b' + 513' -25b + 6 then a is a function of b. A function can be illustrated in our coordinate system as follows. Let us take the equation for frequency versus wavelength as an example. Given different values to the independent variable find the corresponding values of the dependent variable. Then plot the points represented by the different sets of two values. 600 800 1000 1200 1400 1600 1800 2000 500 375 300 250 214 187 167 150 Plotting these points in Figure 31 and drawing a smooth curve through them gives us the curve or graph of the equation. This curve will help us find values of f for other values of 7, (those in between the points calculated) and so a curve of an often -used equation may serve better than a table which always has gaps. When using the coordinate system described so far and when measuring linearly along both axes, there are some definite rules regarding www.americanradiohistory.com HANDBOOK Representation of Functions 785 Figure 32. Only two points are needed to define functions which result in a straight line as shown in this diagram representing Ohm's Law. Figure 33. TYPICAL GRID - VOLTAGE PLATE -CURRENT CHARACTERA the kind of curve we get for any type of equation. In fact, an expert can draw the curve with but a very few plotted points since the equation has told him what kind of curve to expect. First, when the equation can be reduced to the form y = mx + b, where x and y are the variables, it is known as a linear or first degree function and the curve becomes a straight line. (Mathematicians still speak of a "curve" when it has become a straight line.) When the equation is of the second degree, that is, when it contains terms like x' or y' or Ay. the graph belongs to a group of curves, called conic sections. These include the circle, the ellipse, the parabola and the hyperbola. In the example given above, our equation is of the form xy = c, c being equal to 300,000 which is a second degree equation and in this case, the graph is a hyperbola. This type of curve does not lend itself readily for the purpose of calculation except near the middle, because at the ends a very large change in represents a small change in f and vice versa. Before discussing what can be done about this let us look at some other types of curves. Suppose we have a resistance of 2 ohms and we plot the function represented by Ohm's Law: E = 21. Measuring E along the X-axis and amperes along the Y -axis, we plot the necessary points. Since this is a first degree equation, of the form y = mx + b (for E y, m = 2 and / = x and b = 0) it will be a straight line so we need only two points to plot it. - I (line passes through origin) 0 5 E 0 10 The line is shown in Figure 32. It is seen to straight line passing through the origin. be a ISTIC CURVE. The equation represented by such a curve Is so complicated that we do not use it. Data for such a curve is obtained experimentally, and intermediate values can be found with sufficient accuracy from the curve. If the resistance were 4 ohms, we should get the equation E = 4I and this also represents a line which we can plot in the same figure. As we see, this line also passes through the origin but has a different slope. In this illustration the slope defines the resistance and we could make a protractor which would convert the angle into ohms. This fact may seem inconsequential now, but use of this is made in the drawing of loadlines on tube curves. Figure 33 shows a typical, grid -voltage, plate -current static characteristic of a triode. The equation represented by this curve is rather complicated so that we prefer to deal with the curve. Note that this curve extends through the first and second quadrant. Families of curves. It has been explained that curves in a plane can be made to illustrate the relation between to o variables when one of them varies independently. However, what are we going to do when there are three variables and two of them vary independently. It is possible to use three dimensions and three axes but this is not conveniently done. Instead of this we may use a family of curves. We have already illustrated this partly with Ohm's Law. If we wish to make a chart which will show the current through any resistance with any voltage applied across it, we must take the equation E = IR, having three variables. We can now draw one line representing a resistance of 1 ohm, another line representing 2 ohms, another representing 3 ohms, etc., or as many as we wish and the size of our paper will allow. The whole set of lines is then applicable to any case of Ohm's Law falling within the range of the chart. If any two of the three quantities are given, the third can be found. www.americanradiohistory.com . A lt/A ...,4 11 % 1 1 A AA n VA .: 786 ie THE RADIO Radio Mathematics and Calculations IA= AVERAGE PLATE CHARACTERISTICS Et =6.3 v. 1\IA IIÍ III \1 A O II nv3aIM1 IIII/III nvAn..! IiVAV/n /A/'./: M -= I1 III IIIA i 1/4/ . rsr-i,---f MIN011111111111111 AI/AI/I J If e 1/I/I//I/%%/. EEMMIRE o, AN CENZIPMEI , 7 _ e 111111111111 Amaim= 7 F rr NIA nos > N te 111111111111011 AMA ' ft 0 le Figure 34. A FAMILY OF CURVES. fourth variable. But this is not always possible, for among the four variables there should be no more than two independent variables. In our example such a set of lines could represent power in watts; we have drawn only two of these but there could of course be as many as desired. A single point in the plane now indicates the four values of E, I, R, and P which belong together and the knowledge of any two of them will give us the other two by reference to the chart. Another example of a family of curves is the dynamic transfer characteristic or plate family of a tube. Such a chart consists of several curves showing the relation between plate voltage, plate current, and grid bias of a tube. Since we have again three variables, we must show several curves, each curve for a fixed value of one of the variables. It is customary to plot plate voltage along the X-axis, plate current along the Y -axis, and to make different curves for various values of grid bias. Such a e0 M M Atari Jm v0t re0r - v0t r1 JY e7e 00 . Je1 Y < l Figure 35. "PLATE" CURVES FOR A TYPICAL VACUUM TUBE. An equation such as Ohm's Law has three 'variables, but can be represented in Cartesian coordinates by a family of curves such as shown here. If any two quantities are given, the third can be found. Any point in the chart represents o definite value each of E, 1, and R, which will satisfy the equation of Ohm's Low. Values of R not situated on an R line can be found by interpolation. Figure 34 shows such a family of curves to solve Ohm's Law. Any point in the chart represents a definite value each of E, 1, and R which will satisfy the equation. The value of R represented by a point that is not situated on an R line can be found by interpolation. It is even possible to draw on the same chart a second family of curves, representing a NO Y . we have three variables, plate In such voltage, plate current, and grid bias. Each point on a grid bias line corresponds to the plate voltage and plate current represented by its position with respect to the X and Y axes. Those for other values of grid bias may be found by interpolation. The loadline shown in the lower left portion of the chart is explained in the text. of curves is illustrated in Figure 35. Each point in the plane is defined by three values, which belong together, plate voltage, plate current, and grid voltage. Now consider the diagram of a resistance coupled amplifier in Figure 36. Starting with the B- supply voltage, we know that whatever plate current flows must pass through the resistor and will conform to Ohm's Law. The voltage drop across the resistor is subtracted from the plate supply voltage and the remainder is the actual voltage at the plate, the kind that is plotted along the X -axis in Figure 35. We can now plot on the plate family of the set +e Figure 36. PARTIAL DIAGRAM OF A RESISTANCE COUPLED AMPLIFIER. The portion of the supply voltage wasted across the 50,000 -ohm resistor Is represented in Figure 35 as the loadline. www.americanradiohistory.com HANDBOOK tube the loadline, that is the line showing which part of the plate supply voltage is across the resistor and which part across the tube for any value of plate current. In our example, let us suppose the plate resistor is 50,000 ohms. Then, if the plate current were zero, the voltage drop across the resistor would be zero and the full plate supply voltage is across the tube. Our first point of the loadline is E = 250, 1 = 0. Next, suppose, the plate current were 1 ma., then the voltage drop across the resistor would be 50 volts, which would leave for the tube 200 volts. The second point of the loadline is then E = 200, / = 1. We can continue like this but it is unnecessary for we shall find that it is a straight line and two points are sufficient to determine it. This loadline shows at a glance what happens when the grid -bias is changed. Although there are many possible combinations of plate voltage, plate current, and grid bias, we are now restricted to points along this line as long as the 50,000 ohm plate resistor is in use. This line therefore shows the voltage drop across the tube as well as the voltage drop across the load for every value of grid bias. Therefore, if we know how much the grid bias varies, we can calculate the amount of variation in the plate voltage and plate current, the amplification, the power output, and the distortion. Logarithmic Scales Sometimes it is convenient to measure along the axes the logarithm) of our variable quantities. Instead of actually calculating the logarithm, special paper is available with logarithmic scales, that is, the distances measured along the axes are proportional to the logarithms of the numbers marked on them rather than to the numbers themselves. There is semi -logarithmic paper, having logarithmic scales along one axis only, the other scale being linear. We also have full logarithmic paper where both axes carry logarithmic scales. Many curves are greatly simplified and some become straight lines when plotted on this paper. As an example let us take the wavelength frequency relation, charted before on straight cross- section paper. f 300,000 X Taking logarithms: log f = log 300,000 - Logarithmic Scales - log X If we plot log f along the Y -axis and log X along the X -axis, the curve becomes a straight line. Figure 37 illustrates this graph on full logarithmic paper. The graph may be read with the same accuracy at any point in con- 3000 2500 2000 900 900 ,.,,11l'1IIIIII .,,"l'II1IIIIIIIel eaomY 99999999999 1i=91 I999Y9.7MY IIII11rII710 \111111u YYYrY 700 e.e 917 UN 1111111111M=M\1111gR11O1 600 500 IIIMIII1111101111 +0a MEMMIII11111111111111111 -..,'1''t"II'I .,'1111"'ll11' 300 200 11.111111111111111 z 100 8 I . 11 `\11 --..,,,1111''Ill .,,11'lll'lllll 1500 1000 787 --R IIIIIIIIIIIHIIII 8 8 WAVELENGTH IN METERS Figure 37. A LOGARITHMIC CURVE. Many functions become greatly simplified and some become straight lines when plotted to logarithmic scales such as shown in this diagram. Here the frequency versus wavelength curve of Figure 31 has been replotted to conform with logarithmic axes. Note that it is only necessary to calculate two points in order to determine the "curve" since this type of function results in a straight line. tract to the graph made with linear coordinates. This last fact is a great advantage of logarithmic scales in general. It should be clear that if we have a linear scale with 100 small divisions numbered from to 100, and if we are able to read to one tenth of a division, the possible error we can make near 100, way up the scale, is only 1/10th of a percent. But near the beginning of the scale, near 1, one tenth of a division amounts to 10 percent of 1 and we are making a 10 percent error. In any logarithmic scale, our possible error in measurement or reading might be, say 1/32 of an inch which represents a fixed amount of the log depending on the scale used. The net result of adding to the logarithm a fixed quantity, as 0.01, is that the anti -logarithm is multiplied by 1.025, or the error is 21/2%. No matter at what part of the scale the 0.01 is added, the error is always 21/2%. An example of the advantage due to the use www.americanradiohistory.com 1 Radio Mathematics and Calculations 788 10.000 Ï 1.0 0.9 0.8 1000 0.6 0.4 0.3 100 o IS 10 5 0 5 10 15 KC. OFF RESONANCE Figure 38. A RECEIVER RESONANCE CURVE. This curve represents the output of a receiver versus frequency when plotted to linear coordinates. 10 9 8 of semi- logarithmic paper is shown in Figures 38 and 39. A resonance curve, when plotted on linear coordinate paper will look like the curve in Figure 38. Here we have plotted the output of a receiver against frequency while the applied voltage is kept constant. It is the kind of curve a "wobbulator" will show. The curve does not give enough information in this form for one might think that a signal 10 kc. off resonance would not cause any current at all and is tuned out. However, we frequently have off resonance signals which are 1000 times as strong as the desired signal and one cannot read on the graph of Figure 38 how much any signal is attenuated if it is reduced more than about 20 times. In comparison look at the curve of Figure 39. Here the response (the current) is plotted in logarithmic proportion, which allows us to plot clearly how far off resonance a signal has to be to be reduced 100, 1,000, or even 10,000 times. Note that this curve is now "upside down "; it is therefore called a .telectivily curve. The reason that it appears upside down is that the method of measurement is different. In a selectivity curve we plot the increase in signal voltage necessary to cause a standard output off resonance. It is also possible to plot this increase along the Y-axis in decibels; the curve then looks the same although linear paper can 4 3 -20 -lo o +lo +20 KC. OFF RESONANCE Figure 39. A RECEIVER SELECTIVITY CURVE. This curve represents the selectivity of a receiver plotted to logarithmic coordinates for the output, but linear coordinates for frequency. The reason that this curve appears inverted from that of Figure 38 is explained in the text. be used because nuw our unit is logarithmic. An example of full logarithmic paper being used for families of curves is shown in the reactance charts of Figures 40 and 41. An alignment chart con sists of three or more sets of scales which have been so laid out that to solve the formula for which the chart was made, we have but to lay a straight edge along the two given values on any two of the scales, to find the third and unknown value on the third scale. In its simNomograms or Alignment Charts www.americanradiohistory.com ,\.- Figure 40. REACTANCE -FREQUENCY CHART FOR AUDIO FREQUENCIES See text for applications and instructions for use. ` , ' ,. / . 1 Ib. ï/ ' /\ .I4A,1',::^ :`.'::1"-! ,/i I, ''..-.Ìi,Ii,; .I',I .,;-I `:///i% ,\` .//.!1i\.IÌI: \ .... Iyi.. , ' %' ..,. 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THE SIMPLEST FORM OF NOMOGRAM. plest form, it is somewhat like the lines in Figure 42. If the lines a, b, and c are parallel and equidistant, we know from ordinary geometry, that b = 1/2 (a + c). Therefore, if we draw a scale of the same units on all three lines, starting with zero at the bottom, we know that by laying a straight -edge across the chart at any place, it will connect values of a, b, and c, which satisfy the above equation. When any two quantities are known, the third can be found. If, in the same configuration we used logarithmic scales instead of linear scales, the relation of the quantities would become log b = 1/2 (log a + log c) orb = / PNG- ORIGIN o 791 AXIS X Figure 43. THE LOCATION OF A POINT BY POLAR COORDINATES. In the polar coordinate system any point is determined by its distance from the origin and the angle formed by a line drawn from it to the origin and the O-X axis. by the angle A the vectorial angle. We give these data in the following form P = 3 Lo° Polar coordinates are used in radio chiefly for the plotting of directional properties of microphones and antennas. A typical example of such a directional characteristic is shown in Figure 44. The radiation of the antenna represented here is proportional to the distance of the characteristic from the origin for every possible direction. By using different kinds of scales, different units, and different spacings between the scales, charts can be made to solve many kinds of equations. If there are more than three variables it is generally necessary to make a double chart, that is, to make the result from the first chart serve as the given quantity of the second one. Such an example is the chart for the design of coils illustrated in Figure 45. This nomogram is used to convert the inductance in microhenries to physical dimensions of the coil and vice versa. A pin and a straight edge are required. The method is shown under "R. F. Tank Circuit Calculations" later in this chapter. Polar Coordinates Instead of the Cartesian coordinate system there is also another system for defining algebraically the location of a point or line in a plane. In this, the polar coordinate system, a point is determined by its distance from the origin, O, and by the angle it makes with the axis O -X. In Figure 43 the point P is defined by the length of OP, known as the radius vector and Figure 44. THE RADIATION CURVE OF AN ANTENNA. Polar coordinates are used principally in radio work for plotting the directional characteristics of an antenna where the radiation is represented by the distance of the curve from the origin for every Possible direction. www.americanradiohistory.com 792 Radio Mathematics and Calculations Reactance Calculations In audio frequency calculations, an accuracy to better than a few per cent is seldom required, and when dealing with calculations involving inductance, capacitance, resonant frequency, etc., it is much simpler to make use of reactance -frequency charts such as those in figures 40 and 41 rather than to wrestle with a combination of unwieldy formulas. From these charts it is possible to determine the reactance of a condenser or coil if the capacitance or inductance is known, and vice versa. It follows from this that resonance calculations can be made directly from the chart, because resonance simply means that the inductive and capacitive reactances are equal. The capacity required to resonate with a given inductance, or the inductance required to resonate with a given capacity, can be taken directly from the chart. While the chart may look somewhat formidable to one not familiar with charts of this type, its application is really quite simple, and can be learned in a short while. The following example should clarify its interpretation. For instance, following the lines to their intersection, we see that 0.1 hy. and 0.1 pfd. intersect at approximately 1,500 cycles and 1,000 ohms. Thus, the reactance of either the coil or condenser taken alone is about 1000 ohms, and the resonant frequency about 1,500 cycles. To find the reactance of 0.1 hy. at, say, 10,000 cycles, simply follow the inductance line diagonally up towards the upper left till it intersects the horizontal 10,000 kc. line. Following vertically downward from the point of intersection, we see that the reactance at this frequency is about 6000 ohms. To facilitate use of the chart and to avoid errors, simply keep the following in mind: The vertical lines indicate reactance in ohms, the horizontal lines always indicate the frequency, the diagonal lines sloping to the lower right represent inductance, and the diagonal lines sloping toward the lower left indicate capacitance. Also remember that the scale is logarithmic. For instance, the next horizontal line above 1000 cycles is 2000 cycles. Note that there are 9, not 10, divisions between the heavy lines. This also should be kept in mind when interpolating between lines when best possible accuracy is desired; halfway between the line representing 200 cycles and the line representing 300 cycles is not 250 cycles, but approximately 230 cycles. The 250 cycle point is approximately 0.7 of the way between the 200 cycle line and the 300 cycle line, rather than halfway between. Use of the chart need not be limited by the physical boundaries of the chart. For instance, the 10 -µpfd. line can be extended to find where it intersects the 100 -hy. line, the resonant frequency being determined by projecting the intersection horizontally back on to the chart. To determine the reactance, the logarithmic ohms scale must be extended. When winding coils for use in radio receivers and transmit ters, it is desirable to be able to determine in advance the full coil specifications for a given frequency. Likewise, it often is desired to determine how much capacity is required to resonate a given coil so that a suitable condenser can be used. Fortunately, extreme accuracy is not required, except where fixed capacitors are used across the tank coil with no provision for trimming the tank to resonance. Thus, even though it may be necessary to estimate the stray circuit capacity present in shunt with the tank capacity, and to take for granted the likelihood of a small error when using a chart instead of the formula upon which the chart was based. the results will be sufficiently accurate in most cases, and in any case give a reasonably close point from which to start "pruning." The inductance required to resonate with a certain capacitance is given in the chart in figure 41. By means of the r.f. chart , the inductance of the coil can be determined, or the capacitance determined if the inductance is known. When making calculations, be Tank Circuit Colculotions R. F. sure to allow for stray circuit capacity, such as tube interelectrode capacity, wiring, sockets, etc. This will normally run from 5 to 25 micro microfarads, depending upon the components and circuit. To convert the inductance in microhenries to physical dimensions of the coil, or vice versa, the nomograph chart in figure 45 is used. A pin and a straightedge are required. The inductance of a coil is found as follows: The straightedge is placed from the correct point on the turns column to the correct point on the diameter -to- length ratio column, the latter simply being the diameter divided by the length. Place the pin at the point on the plot axis column where the straightedge crosses it. From this point lay the straightedge to the correct point on the diameter column. The point where the straightedge intersects the inductance column will give the inductance of the coil. From the chart, we see that a 30 turn coil having a diameter -to- length ratio of 0.7 and a diameter of 1 inch has an inductance of approximately 12 microhenries. Likewise any one of the four factors may be determined if the other three are known. For instance, to determine the number of turns when the desired in- www.americanradiohistory.com COIL CALCULATOR NOMOGRAPH Figure 45. For single loyer solenoid coils, any wire size. See N PLOT OF AXIS TURNS - - text for instructions. RATIO DIAMETER LENGTH 865- 400 - 20000 300 4 -10000 - 200 150 -100 - INDUCTANCE IN MICROHENRIES 90 80 70 80 50 -- -- -4 4000 3000 2- -3 1000 800 600 400 300 -2 200 -- -S 3- - 2000 -40 8000 6000 - DIAMETER INCHES 100 80 60 40 30 20 .8- -- - -- -1.5 .8- 1Ó 8 - r- 6 20 --4 3 -15 - 10 2 - .6 -.75 -.4 -.3 -.2 -5 -- o --4 3 www.americanradiohistory.com 794 Radio Mathematics and Calculations D/L ratio, and the diameter are known, simply work backwards from the example given. In all cases, remember that the straightedge reads either turns and D/L ratio, or it reads inductance and diameter. It can read no other combination. The actual wire size has negligible effect upon the calculations for commonly used wire sizes (no. 10 to no. 30). The number of turns of insulated wire that can be wound per inch (solid) will be found in a copper wire table. ductance, the Significant Figures In most radio calculations, numbers represent quantities which were obtained by measurement. Since no measurement gives absolute accuracy, such quantities are only approximate and their value is given only to a few significant figures. In calculations, these limitations must be kept in mind and one should not finish for instance with a result expressed in more significant figures than the given quantities at the beginning. This would imply a greater accuracy than actually was obtained and is therefore misleading, if not ridiculous. An example may make this clear. Many ammeters and voltmeters do not give results to closer than 1/4 ampere or 1/4 volt. Thus if we have 21/4 amperes flowing in a d.c. circuit at 63/4 volts, we can obtain a theoretical answer by multiplying 2.25 by 6.75 to get 15.1875 watts. But it is misleading to express the answer down to a ten -thousandth of a watt when the original measurements were only good to 1/4 ampere or volt. The answer should be expressed as 15 watts, not even 15.0 watts. If we assume a possible error of 1/8 volt or ampere (that is, that our original data are only correct to the nearest 1/4 volt or ampere) the true power lies between 14.078 (product of 21/8 and 65/8) and 16.328 (product of 23/8 and 67/8). Therefore, any third significant figure would be misleading as implying an accuracy which we do not have. Conversely, there is also no point to calculating the value of a part down to 5 or 6 significant figures when the actual part to be used cannot be measured to better than 1 part in one hundred. For instance, if we are going to use 1% resistors in some circuit, such as an ohmmeter, there is no need to calculate the value of such a resistor to 5 places, such as 1262.5 ohrr>i. Obviously, 1% of this quantity is over 12 ohms and the value should simply be written as 1260 ohms. There is a definite technique in handling these approximate figures. When giving values obtained by measurement, no more figures are given than the accuracy of the measurement permits. Thus, if the measurement is good to two places, we would write, for instance, 6.9 which would mean that the true value is somewhere between 6.85 and 6.95. If the measurement is known to three significant figures, we might write 6.90 which means that the true value is somewhere between 6.895 and 6.905. In dealing with approximate quantities, the added cipher at the right of the decimal point has a meaning. There is unfortunately no standardized system of writing approximate figures with many ciphers to the left of the decimal point. 69000 does not necessarily mean that the quantity is known to 5 significant figures. Some indicate the accuracy by writing 69 x 10' or 690 x 10' etc., but this system is not universally employed. The reader can use his own system, but whatever notation is used, the number of significant figures should be kept in mind. Working with approximate figures, one may obtain an idea of the influence of the doubtful figures by marking all of them, and products or sums derived from them. In the following example, the doubtful figures have been underlined. 603 34.6 0.120 637.720 answer: 638 Multiplication 654 0.341 654 0.342 19612 26116 1308 2616 1962 223.668 1108 answer: 224 224 It is recommended that the system at the right be used and that the figures to the right of the vertical line be omitted or guessed so as to save labor. Here the partial products are written in the reverse order, the most important ones first. In division, labor can be saved when after each digit of the quotient is obtained, one figure of the divisor be dropped. Example: www.americanradiohistory.com 1.28 527 1 673 527 53 146 106 5 40 40 INDEX 795 www.americanradiohistory.com A 414 92 694 734 Absorption, Ionospheric Acceptor, def. of A.C. -D.C. power supply A -C V -T voltmeter Adaptor, F -M Admittance, def. of "A" -frame antenna mast Air capacitor Air dielectric Air -gap, capacitor B.f.o. Filter Alignment " - - ---- Receiver I -F All- driven antenna Alpha, def. of Alternating current " 444 355 32 262 Amplitude Effective value Def. of Generation - Amateur frequency bands Amateur Radio Ammeter Ampere, def. of " - D.C. " " " 47 45, 46 45 61 59 52 42 12 11 731 22, 23 36, 62 73, 74 678 225, 212 169 168 27, 256 127 108, 118 108, 137 129, 249 124, 249 118, 123 117 613 296 126 110, 111 129 1 99, AB1 99, 108, 123, 108, 113, inductive distortion products " 93 45 Average 41, 45 42 -- coupling " " " " " turns Class AB Class B Class C " 500 Amplification factor Amplifier audio, 15 -watt " Audio Cascode cascode grounded -grid constant current curves -- " " licenses Cathode driven Cathode Cathode follower A2 Class A Response R.F., Class AB1 R.F., Class B 234 233, 234, 180 Impedance "J" operator Phase angle Ohm's law R.M.S. Pulsating Transformer Transient Voltage divider Alternator Push -pull " " 321 52 R -F " Doherty Driver Equivalent circuit feedback 121 Grid circuit 162 Grounded cathode, grounded screen 162, 212, 256, 612 Grounded -grid 612 Grounded -grid, use of 617 350 w., grounded -grid 627 813 grounded -grid 634 KW -2 grounded -grid 146 Hi -fi, 25 -watt 138, 112 High mu (A) Hi -fi 171, 208 I -F Horizontal 649 Kilowatt 649 Kilowatt 4 -400A 284 Linear 654 Linear, 2kw. P.E.P. 166 Load line 118 Loftin -White 250, 615 Neutralization 152 Noise factor 199 Operational 120 Pentode 609 Pi- network 159 Plate efficiency 18, 602 Power " " " -F R -F 112 166 168 611, 612 Grounded screen RC 109 4CX -1000A 211 Semi -conductor 104 200 201 296 Summation Summing Terman- Woodyard Tetrode Transistor 606 104 "Tri- band" 4CX300A Ultra- linear Vertical Video V.H.F. 622 s 142 138 112, 113, 128 "strip -line" " Voltage " Williamson " 3 -10002 " 4 -400A all -band " 4CX1000A, 2 kw. Amplitude A.C. Amplitude, Distortion Modulation..109, Analog computers problems "And -or" circuitry Angle of radiation V.h.f. 407, Anode, def. of dissipation Antenna Adjustment " All- driven " Bandwidth Beam design chart 2- element Bidirectional Bi- square Bobtail Broadside - - --- - - - Bruce Center -fed Colinear Combinations of Construction Corner reflector Couplers mobile - Coupling systems Cubical quad Curtain Delta match Dipole array Directivity - broad -band Discone Doublet Dummy Efficiency Element spacing - - - 1 -- R.F., R 121, 604 End -effect End -fed end -fire Feed systems Franklin array Fuchs Gain, beam 140 661 643 654 45 280, 669 195, 200 204 456, 474 67, 72 505 500 401, 409 494, 491 501, 466 468 464 466 434 462 471 444, 502 481 450, 516, 520 447 467 464 497 461, 431 400, 406 437, 477 423 736 405 464 402 422, 433, 469 424, 494 462 422 491 499 405 427 422 455 404 473 Lazy -H 464 length Length -to- diameter ratio_.401, 402 Long wire 457 Gamma t. 595 110 match Ground loss Ground plane Hertz High frequency type Impedance Intercept, v.h.f. - 796 www.americanradiohistory.com " " " " " " " " " " " " to vt It 1. Marconi Matching systems Measurements Multee Multi -band Mutual coupling Parasitic beam 404, 428 438 740 436 432, 510 475 490 400, 456 462 400 475 Patterns Phasing Polarization Polarization, v.h.f. Power gain 0 401 - 405 404, 400 460 490, 498 Reactance Resonance Rhombic Rotary Rotary match - Rotator Single wire feed Six- shooter Sleeve Space conserving Stacked Sterba Stub adjustment T -match 3- element beam 508 437 468 476 - stub match 430 464, 494 465 441, 499 497 484 tt Triplex 471 to Tuner 452 476 434 458 426 to ft el 0. O. OS tt to Turnstile Two -band Marconi V-type Vertical V.h.f. 8 element, ground plane....484, 479, V.h.f. helical horn V.h.f. nondirectional V.h.f. rhombic 487, V.h.f. Yagi Screen array W8JK X -array -- - - Yoke match t. "Zepplin" Antennascope Antennascope, SWR meter Anti- resonance Aquadag Area, capacitor plate Arrays, antenna "Assumed" voltage atomic number Atom, def. of - - Attenuation, Bass /Treble Amplifier Audio " 15 watt " Distortion Equalizer Feedback loop Filter, SSB Frequency, def. of Hum limiters Phase Inverter - SSB Phasing network Rectification Wiring technique Autodyne detector Automatic load control, SSB Automatic modulation control Automatic volume control Autotransformer Avalanche voltage 477 482 478 482 484 469 465 494 423, 463 748 747 54 85 33 461, 408 52 21 139 225 678 135 139 141 329 42 142 155, 185, 226 115 334 379 142 206 341 670 223 63, 386 697 - B Balanced modulator SSB 327, 334, 339 350 Balanced modulator, deflection 746 Balanced SWR bridge 744 Balanced transmission line measurements 12 Bands, amateur 215 Bandspread tuning 401, 409 Bandwidth, antenna Bandwidth, modulation Base electrode Bass suppression, audio Battery bias Beam deflection modulator " Design chart Dimensions " Element spacing Front-back ratio H -F 5- element 2- element - - - - Power tube - 492 206 222 36 -H curve Bias " 267 266 Battery Cathode Def. of cut -off Grid Grid leak R - -F 73 108, 265 112, 266 604 266 305, 307 675 amplifier Safety Shift modulator Supply, modulator Supply, regulated Transistor Bidirectional antenna Billboard antenna Binary notation Bishop noise limiter Bi- square antenna Bistable multivibrator Bit 71 2 96 501 473 195, 196 226 466 102 196 376 27, 709, 391 394 397 102, 189, 190 468 70 139 203 737 738 716 689, 690 738 - Blanketing interference Bleeder resistor safety Blocked grid keying Blocking diodes Blocking oscillator Bob -tail antenna Bombardment, cathode Boost, Bass /treble Break voltage Bridge, impedance " Measurements " Power supply " Rectifier " Slide -wire " 92 304 267 350 494, 492 493 492 492, 490 78 491 Radiation resistance " Three element HF 220 Mc. Beat note Beat oscillator B 281 454, 746 SWR Bridge -T oscillator Bridge -type vacuum tube voltmeter Bridge, Wheatstone Broad band dipole Broadcast interference Broadside antennas arrays Bruce antenna Butterfly circuit - - C Calorimeter Capacitance Calculation " Interelectrode " Neutralization " Tank Stray 797 www.americanradiohistory.com - definition 192 131 738 431 375 464, 408 466 231 736 32, 30 76, 106 107 259, 216 268 46 Capacitive coupling Capacitive reactance A.C. circuit Capacitor " Air Characteristics " Breakdown Charge Color code Electrolytic Equalizing Filter -- Clipping, negative peak Clipping, phase shift Clipping, speech 34 355, 354 Closed loop feedback 262 Coaxial reflectometer Coaxial transmission line Coaxial tuned circuit 31 527 34, 708 Cade Code, 34 707, 690 30 Fixed Input filter Leakage Low inductance Parallel 714, 690 Coercive force Coil, core loss Coil, Q 34 611 33 55 33 32 39 Q Series Colinear antenna Collector electrode Collins feed system Collins filter Color code, component Color code, standard Colpitts oscillator Complementary symmetry Complex quantity Component color code Component nomenclature graph Compound, def. of Temperature coefficient Time constant Vacuum variable vacuum 360 Voltage rating 34 Distortion Carrier 309 282 Carrier, Shift Carrier, S.S.B. 323, 327, 328, 330 V.H.F. 111, 212 Cascade amplifier Bias Cathode 108, 266 115, 116 " Coupling coupled inverter Definition " Current 79, 67 " Driven amplifier 612, 256 Follower 272 Follower amplifier 127, 164 Follower driver 680, 683 Follower modulator 288 Keying 393 Modulation 295 Ray oscilloscope tube 170, 84 Tank, grounded -grid 165 Cavity, resonant 230 Cell, Weston 24 Center -fed antenna 434 Ceramic dielectric 32 Charge 22, 23, 30 - - -- Compression, volume Compressor, A.L.C. Computers, electronic Conductance, def. of Conduction Conductor Conductivity Constant amplitude recording Constant current curve Constant -K filter Constant velocity recording Constants, vacuum tube Control circuit, mobile Control circuit, transmitter Conversion conductance Conversion frequency, SSB Converter stage Converter, transistor - - Chassis layout Choke, filter Choke input filter Choke, R -F Circuit constants, measurement of Circuits Coupled " D.C. Equivalent " Input loading " Limiting Magnetic Parallel Parasitic -- Q 1. Resonant - Series Circulating current Clamping circuit Clapp oscillator Class Class Class Class Class Class Class Class Class Class amplifier, def. of A2 amplifier AB amplifier, def. of ABI amplifiers B amplifiers B amplifier, grounded -grid B modulator C R -F amplifier amplifiers C bias B, Clipper -Amplifier design Clipper filter, speech Clipping circuits 726 709 713, 715 270, 357 737 Core loss Corner reflector antenna Corona static Coulomb Counter e.m.f Counting circuit Capacitive Coupling " Cathode Choke Critical Direct Effect of Impedance Inductive Interstage r -f Link 153 21, 51 152 185 - 35 25 361 55 -- Tank ....53, 24, 55, 57 56 187, 188 A Magnetic 239 108 118 108 108 108, 123, 125, 159 613 126, 292 249 108, 124, 125, 249 267 678 302 practice set Coefficient of coupling Coefficient, temperature Parasitic R -F feedback Systems, antenna Transformer 670 299 298, 670 192 742 419 230 14 -20 20 38 32 37 55 54, 356 462 92 443 220 528 527, 528 239 97 49 528 526 21 670 341 194 52 67, 90 23 137 74, 157, 168 64 136 107 521 387 80 336 209 100 55 481 525 22, 31 37 190 268 115 114 57, 216 115 56 113, 269 268 270, 603 38 360 273 447 113 Unity Critical coupling Critical inductance Cross modulation Crossover point 269 216 686 377 136 C.R.P.L. Predictions 414 244 218, 332 244 Crystal " " 185 -- Current Filter Lattice filter Harmonic 798 www.americanradiohistory.com - " Oscillator Quartz " Pickup Cubical quad antenna Alternating Current " Amplification (alpha) " Cathode Circulating Closed path Def. of " Effective Effective (a.c.) - " " " " 93 79 - 56, 28 106 42, 44 693 731, 733 125 amplifier - " Rating, Power supply Saturation Skin effect Curtain antenna " 414 D D. C. clamping circuit D.C. power supplies D.C. restorer circuit Defector autodyne Deflection, plate Degeneration, transistor Degree, electrical Delta match antenna Demodulator D. C. -- - - 187 684 188 206 crystal - -- 96 43 system....497, 426, 438 205 - - 1 329, 349 256 258 423, 425 93 126 680, 683 411 726 95 522 240 E 171 208 Detection, slope Detection, synschronous, DSB 349 222 Detector Diode 208 " Fremodyne 222 Grid leak Impedance Plate Product 235, 348 Ratio 319 207 Super -regenerative Deviation, FM Measurement 310, 316 Dielectric, ceramic Constant (K) 30, 31, 32 Differential keying 396 198 Differentiation, electronic Differentiator (RC) 59 Digital circuits 195 computers Digital package 204 Diode A.v.c. 223 " Blocking 397 232 Crystal Def. of 71 Detector 222 1. 204 Gate Limiter 85 Mixer 210 Modulator 328 90 Semi -conductor 103 Storage time Voltmeter 734 105 Zener Dipoles 399, 432, 461, 494 Direct current circuits 21 455 Directive antennas, H -F Directivity, antenna 400, 406 30 Discharge of capacitor Discone antenna 437, 477 421 Discontinuities, transmission line 318 Discriminator, F -M - 98 190 26 296 334 213 - 731 117 amplifier 109, 135 613 340 135 - 41 d'Arsonval meter 309 119 136 305 Transient Transistor Divider, frequency Divider, voltage Doherty amplifier Dome audio phasing network Double conversion circuit Double sideband, DSB Doubler, frequency Doubler, push -pull multi -wire Doublet, antenna Drift transistor Driver, amplifier Driver, cathode follower Duct propagation Dummy antenna dummy loads Dynamic resistance Dynamotor, PE -103 Dynatron oscillator 93 73 285 64 Cycle Cycle, sunspot Frequency Products Products, SSB 685 72, 55 464 Cutoff (alpha) Cutoff bias Cutoff, extended Cutoff frequency 72 165 109 135 - 22 733, 45 Electrode Induced Instantaneous Inverse Measurement - - 41 - Peak Dissipation, anode Dissipation, grid Distortion Amplitude " Audio " Carrier " Harmonic Intermodulation Modulation Phase Nonlinear 242, 243 137 467 38 Eddy current Efficiency, antenna Electric filters Electrical energy Electrical potential 405 63 29 22 Electromagnetic deflection Electrolytic capacitor Electrolytic conductor Electromagnetism Electromotive force (e.m.f.) Electron coupled oscillator drift orbit Electron, def. of Electronic computers Electronic conduction integration Electronic differentiation Electronic Keyer, "910" Electronic multiplication Electrostatics Electrostatic deflection energy metallic, non -metallic Element, def. of Element, reactive, non- reactive Emission equation photoelectric Emission " Secondary " Spurious " Thermionic Emitter electrode - - - - Enclosures End effect, antenna End -fed antenna End -fire antennas - -- arrays Electrostatic Electrical Energy " Potential " Storage, capacitor " Transistor ENIAC Envelope, carrier Equalizer, audio Equal- tempered scale Equivalent circuit 799 www.americanradiohistory.com 86 34, 708 22, 67 35 22, 37 239 67, 21 194 67 198 597 198 30 83, 30 21, 90 21 70 67 71, 77 379 67 92 724, 725 402 422, 433 469, 408 29, 30 30 31 93 195 280 139 135 51, 110, 111 Equivalent circuit, transistor Equivalent noise resistance Error signal Excitation, grid Exciters, low power Exciters, SSB Extended -Zepp antenna Pulse repetition 95 153 193 250 576 342, 345 463 Shift keying Sound Spectrum Spotter Standard WWV Sweep Frequency Modulation F Factor of merit Fading Farad, def. of Feedback amplifier 54 414 Audio 141 " " Circuits Control Error cancellation 192 192 193 199 " " " " 31 Miller Transistor Feed systems, antenna Feedthrough power Field, magnetic Filament, def. of Filament reactivation Filter Capacitor " Capacitor input " Carrier, SSB " Choke Choke input Circuits, mobile Crystal Crystal lattice Generator, SSB High Pass Inductor input Insertion loss 101 - 713, 715 514 SSB 218 332 330 64, 368 687 65 64, 372 M- derived 64 Mechanical 220, 332 Noise 225 Passband, SSB 332 O- multiplier 234 Resistance- capacitance-Resonance -.687, 688 Ripple factor 688 Sections Series Shunt 64 TVI, receiver TVI -type 65, 372 Wave 63 Filter -type exciter, SSB 345 Fixed bias 108 Flat -top beam antenna 469 Fletcher- Munson curve 140 Floating Paraphase inverter 116 Flux, def. of density 35, 36 Flywheel effect 57, 257 Folded dipole 440, 425, 494 Forcing function 200 Foster -Seely discriminator 319 Franklin antenna 462 Franklin oscillator 241 Free electrons 21 Free -running multivibrator 189 Fremodyne detector 208 Frequency 41, 58 " Conversion, SSB 336 " Cutoff 64 " Distortion 109 Divider 190 -- - - Multiplier 207 413 739 256 321 index 310 318 320 314 311 321 317 492 422 antenna Full -wave limiter Full -wave rectifier 612 35 67 68 707, 690 714, 690 330 709 low pass Interruption Maximum useable Measurements Fuchs 424, 494 - - - 740 739 172 308 Reception Front -back ratio, beam 272, 607 R -F " Adapter 41 Deviation ratio Discriminator Limiter Linearity Narrow band Pre -emphasis 129 " 190 136 176 53 322 134 Range, hi- fidelity Ratio, Lissajous Resonant Function, Forcing Function generator 228 - non -linear - 689 200 ramp --202, 203 G -- - -- G, conductance, def. of Gain, antenna beam Gain, power resistance voltage Gamma match, antenna system Gas tube generator Gate, diode limiter Gauss, def. of Generator, function Generator noise 52 491, 457 94 499, 440 87, 172 204, 185 36 202 524, 750 Generator, sawtooth 172 Generator, time base 171 Gilbert, def. of 36 Grid bias 108, 265 Grid, def. of 72 Grid dissipation 165 Grid excitation 250 Grid leak bias 109, 112, 266 Grid leak detector 222 Grid limiter, limiting 187 Grid modulation 250, 286 Grid neutralization 251 Grid-screen factor 78 Ground bus 142 Ground currents 358 Ground loss, antenna 405 Ground plane antenna VHF 427, 477 Ground resistance 406 Ground, R -F 358 Ground termination 429 Ground, transmitter 390 Ground, wave 410 Grounded -cathode amplifier 162 Grounded -grid amplifier 162, 212, 256, 612 Grounded -grid cascade amplifier 169 Grounded -grid, cathode tank 165 Grounded -grid r.f. amplifier 612 Guy wires, antenna 445 - H Hairpin coupling Half -wave rectifier Harmonic B.f.o. " Crystal " Def. of - 800 www.americanradiohistory.com 230 689 223 244 58 " " " " Distortion 119 Initial condition voltage 203 Music 135 Injection voltage Input loading Input resistance Insertion loss, filter 211 Oscillator Radiation Harmoniker TVI filter Hartley oscillator Hash rectifier 245 260, 369 375 238 694 353 99 104, 707 70 292 479 37 422 206 455 134 -138 382 368 Heat cycle, resistor Heat sink Heat sink, transistor Heater cathode Heising modulation Helical antenna Henry, def. of Hertz antenna Heterodyne H -F antennas High fidelity High fidelity, Interference High -pass TVI filter Holes in semi -conductor 91 Horizontal directivity Horn antenna, VHF Hot cathode phase inverter Hum, audio Hysteresis loop loss 406 482 116 - 142 36, 38 I -f alignment 180, 233 " Amplifier 208 " Noise limiter 225 " Pass -bond 217 Rejection notch 220 " Shape factor 217 " Tuned circuit 216 Ignition noise 523 Images -image interference- ratio....209, 211, 381 Impedance 46, 47 " Antenna 404 " Bridge 737 " Complex 51 " Coupling 56, 113 " Match, audio 127 " Reflected 56, 63 " Resonant 54 " Screen circuit 287 I " " 260 Transformation 63 97 " Transistor " Transmission line 417 " Triangle 48 742 Incident voltage, transmission line 748 Indicator, Antennascope Indicator, Selsyn 510 Indicator standing wave twin- lamp..741, 745 Induced current 42 37 Inductance " Capacitor 355 " Cathode lead 153 " Critical 686 " Lead Magnetic Mutual 81, 37 " Parallel Series 38 " Resistor 353 " Screen lead 255 Induction, def. of 42 Inductive reactance 46 Inductive coupling 269 Inductive tuning 609 Inductor, iron core 38 Inductor, time constant 40 Infinite impedance detector 222 " - - -- Instability, R -F amplifier Instability, static Insulation, VHF Insulator, def. of Integration amplifier Integration, electronic Integrator, Miller Integrator (RC) - 607 117 475 22, 23 200 198 199 - - Interelectrode capacitance Interference Broadcast " Harmonic " Hi -fi image TV Interlock, power Intermodulation distortion Intermodulation test Internal resistance International Morse Code Interruption frequency Interstage coupling Intrinsic semi -conductor Inverse current Voltage Inverter, phase voltage divider Ion, def. of Positive Ionosphere, absorption Ionosphere, Sporadic -E layer Ionospheric cycle Ionospheric fading Ionospheric layers propagation Ionospheric reflection IR drop - -- - 59 76 375 369 381, 382, 367 391 136 145 25 15 207 268 91 115, 1 693 16, 117 22, 87 13, 414 414 414 414 412, 413 415 24 Iron vane meter 733 Isotropic radiator 406 J Jitter Johnson 'J" 0 -feed 188 442 system Operator 47 Joule, def. of Junction, transistor 30, 37 93 K 420 Surge Tank circuit 152 107, 214, 274 65 dielectric constant 32 Key clicks Keyer, "9T0" electronic Keying blocked grid Cathode circuit 392 K, - " 597 394 393 396 322 399 395 42 Differential Frequency shift Screen grid Transmitter Kilocycle Kinescope tube Kirchhoff's Laws Kylstron reflex 84 - 28 81, 82 L Lamb noise limiter Layout, chassis Lazy -H antenna L/C ratio Lead inductance Leakage, capacitor Leakage reactance Left -hand rule Length, antenna L- section filter 801 www.americanradiohistory.com 226 726 464 56 81 34 63 35 401 64 audio diode 226 Mechanical filter, use of Megacycle 185 Megohm -M full -wave noise series diode 320 228 225 Memory circuit Mercury vapor rectifier Meteor bursts Meter, d'Arsonval Meter, iron vane Meter, multi -range Meter, rectifier Meter shunt Meter, thermocouple Mho, def. of Mica dielectric Microfarad Microhenry Micro -ohm Middle C Miller effect Miller feedback oscillator Miller integrator Milliammeter Millihenry, def. of Mixer circuits " Diode " Noise " Products, SSB spurious 12 Licenses, amateur Limiter, Limiter, Limiter, Limiter, Limiter, Limiter, Limiter, Limiting Limiting F 185 228 185, 186, 202 204 INS circuit diode Line filter Line regulation Line, Slotted Linear amplifier " Class " Kilowatt B " " Tuning 2KW, P.E.P. Linear matching transformer Linearity, distortion products Linearity tracer antenna Link coupling Lissajous figures Litz, wire L- network design Load line Load line, r.f. amplifier Load line, transistor Load resistance amplifier Load, r.f. dummy loaded -O Loftin -White amplifier Long -wire antenna Loop feedback Loran, band Loudness control, audio response Loudspeaker def. of Low- frequency parasitics Low -pass TVI filter - - - 227 384 740 284 129 649 285 654 442 613 185 270, 448 176 55 263 74 166 99 119 736 57 118 457 192 12 140 139, 140 362 372 M -- - Magic eye tube Air gap Magnetic field flux " Circuit " Eddy current " Hysteresis loss " Induction " Left hand rule reluctance " Permeability residual Magnetism Magnetomotive force (M. M. F.) - - Magnetron Magnitude, scalar Majority carrier Marconi antenna Mast, A -frame Matching stub, antenna Matching systems antenna Matching transformer Mathematics, radio Maximum usable frequency Maxwell, def. of - M- derived filter Antenna Measurements Bridge Circuit constants Voltage Current Frequency Parallel wire line Power Transmission line Mechanical filter 88, 225 38 35 38 38 - 204 694 416 731 733 732 734 731 734, 736 74 32 30 37 23 134 107, 112, 218 199, 245 199 731 37 210 80 210 - - - " SSB " Stage " Transistor Triode " " 337, 339 336 209 100 211 Tube 79 Mobile, Antenna coupling " " Control circuit Dynamotor Equipment construction Filter circuits Noise limiter Noise sources Power supply Mode of resonance 516, 520 - 521 522 design -...520, 511 Modulation " Amplitude Automatic control Bandwidth Cathode 37 Class 35 Constant efficiency Distortion Frequency Grid Heising Index, F -M Pattern, oscilloscope Percentage saturation....36 35, 37 36 83 48 93 408, 428 B Suppressor Transformer Variable efficiency -- Velocity Modulator " 740 738 737 731, 733 739 745 " " Adjustment Balanced S.S.B. Beam deflection Bias -shift Cathode follower Class B Construction Crosby Diode Matching impedance match 731,735 - 740, 744 220 514 512 525 515,517,697 230 237 280, 669 670 281 295 292 284 305 308 250, 286 292 310 178 282 311,315 Phase Plate Screen 444 441, 499 438 425 752 13, 413 36 64 571 42 24 802 www.americanradiohistory.com 249, 291, 293 286 290 293 283 81 676 328, 334, 339 350 305 288 125 672 590 328 125, 127 " Reactance tube " " Semi -conductor 312 104 336 669 683 697 SSB " Tetrode Zero bias " 200 -watt Molecule Monitor oscilloscope Morse code " 21 179 15 Mu factor (µ) 78 Multee antenna Multi -band antenna Multiplication, electronic Multiplier, frequency Multiplier resistor Multi -range meters Multivibrator circuits Multivibrator, free running Multi -wire doublet Music, def. of Music systems scale Mutual conductance Mutual coupling Mutual coupling, antennas Mutual inductance Mycalex, dielectric - 436 432, 510 198 256 732 732 102, 188 189 Ohm's Law (complex quantities) Ohm's law for magnetic circuit Ohm's Law (Resonant Circuit) Ohms - per -volt Omega, def. of One -shot multivibrator On -off circuits Open wire line Operating desk, constuction of Operational amplifier Optimum working frequency Orbital electron Oscillation, parasitic Oscillator Beat " " 425, 439 134 138, 135 73 - Procedure R -F Shunt Test Octave, def. of Oersted, def. of Ohm Ohm's Law Ohmmeter, low range Ohm's Law (a.c.) - 241 173 245 238 247 245 192 247 670 Pierce 245 199 263 250 107 RC 191 Relaxation T.p.t.g. Transistor Transistron V.F.O. 102 239 251 615 277 274 252 276 100 - 19 242 240 239 NBS bridge -T Overtone Phase shift 253 Generator, silicon Limiters Mobile Mixer Regulator sources Suppression Thermal agitation Voltage Wow and flutter Nomenclature, color code Nomenclature, component Nonconductor Nondirectional VHF antenna Non -linear distortion Non-linear function Non -resonant transmission line Nonsinusoidal wave N -P -N transistor "Nuvistor" tube, use 21 277, 361, 608, 615 222 102, 189, 190 311 192 141 Transistor - 199 414 239 Miller Neutralizing procedure NI (ampere turns) Noise Factor " Check, mobile - Franklin Free running Harmonic Hartley Keying 471 37 32 417 724 192 248 E.c.o. 216 N Narrow -band F -M NBS bridge -T oscillator Negative feedback loop Negative peak clipping Network, R -L integrator Networks, L and Pi Neutralization Amplifier Capacitance " Grid " Grounded -grid - Blocking Bridge -T Circuits Clapp Colpitts Code practice Crystal Dynatron 49 36 54 732 44 189 195 62 152 524 749 225, 512 210 524 225, 523 191 101 241 242 Wein- bridge 191 Oscilloscope 170 " Circuit 174 " Linearity tracer 183 " Lissajous figures 176 Modulation pattern -phase patterns.178, 177 1. Monitor 179, 181, 750 Sideband measurements 182 Trapezoidal pattern -wave pattern..178, 180 peak Output, power 124 Overloading, TV receiver 367 Overtone crystals 247 Overtone, music 135 Oxide filament 67, 69 188 151 - 135 P Paper dielectric Parallel circuit resistance " Diode limiter 527 526 22 478 135 202 417 Feed 271 " Resonance 54 745 Wire line measurements Parasitic antenna, Design chart " Antenna, VHF " 547 " Beam design Check for " Coupling Element, antenna o Oscillation 135 36 Resonance band, I -F band, mechanical filter band, SSB filters Patterns, antenna Pass Pass Pass 23, 24 733 45 25 186 " " 58 92 32 803 www.americanradiohistory.com 494 484 490 364 360 493 277, 361, 608 360 217 221 332 400, 456 PE -103 Peak Peak Peak Peak Peak Peak 522 dynamotor amplifier current current (a.c.) 1 envelope power, limiter SSB noise limiter power output Peaked wave Pentode amplifier Pentode tube Period, sound Permeability Phantom signal angle (a.c.) Phase " Angle difference " Distortion " Inverter Modulation Shift, clipping Shift, feedback Shift, oscillator Phasing, antenna Phasing generator, SSB Phasing networks, audio Phonograph reproduction Photoelectric emission - - Pickups Pickup, spurious Pierce oscillator - Pi- network amplifier design Pi- network chart Pi- network coupling Pilot carrier, SSB Pitch, sound Plate current flow Plate current, static Plate detector Plate efficiency amplifier Plate modulation Plate P -N -P Point Polar resistance transistor 45 324 185 225 124 59 120 77 134 36 378 46 177, 178 109, 135 115 311, 315 299 193 191 462 333 334 136 67 137 380 245 609 265, 263 449 323 134 257 166 222 159 249, 291 73, 74 91 contact transistor - notation Polarization, antenna VHF Polyphase rectifier Potential difference Potential, electrode Potential energy Power amplifier design Power amplifier triode Power, feed through Power, gain Power gain antenna Power gain SSB Power measurement Power interlock Power -line filter Power, resistive Power supplies " A.C. -D.C. Bridge Design Dual voltage Mobile Oscilloscope Regulated Screen Three -phase 1. 25 Transistorized Voltage doubler Voltage multiplier Voltage quadrupler 92 48 400, 475 693 22 106 30 602 118 612 94 401 324 731, 735 " 61 133 235, 348 409 -416 414 Propagation Propagation, sporadic -E Pulsating alternating current Pulse- repetition frequency Push -pull amplifier Push -pull transformer Push -pull tripler doubler Push -to -talk circuit 45 103, 190 604, 121 - 113 258 522 Q 158 405 356 Q amplifier tank Q antenna Q coils 55 Q, def. of Q loaded 57 Q multiplier Q tank circuit Q transformer Q tuned circuit 234 259 425 214 Quad antenna Quadrant, sine Quantity, complex Quench oscillator 467 43 49 207 R 43 Radian, def. of Radiation, angle of Radiation, def. of Radiation, harmonic Radiation pattern, distortion Radiation resistance Radiator, isotropic Radiator cross- section, VHF Radio frequency, def. of Radio mathematics Radio propagation Radio teletype Ramp function Ratio detector Ratio, image amplifier RC audio network 391 RC differentialor 225 RC oscillator 29 RC time constant 684 694 716 713 716, 718 697 173 712 267 RC 703 695 695 695 138 137 212 Preselector Primary transformer Probe, R.F. (v.t.v.m.) Product detector RC 517 716 717 718 383 709 99 386 300 volt 1500 volt " 2500 volt Power system, primary Power transformer Power transistor Powerstat auto- transformer Preamplifier, hi -fi Pre -emphasis, recording " -- 407, 456, 474 399 369 474 400, 491 406 475 42 752 409 322 203 319 211 integrator circuits transient Reactance, antenna Reactance, capacitive Reactance, leakage - 191 inductive Reactance, resonance Reactance tube modulator " Broadcast transistorized " DX operator Mobile transceiver, 10 804 www.americanradiohistory.com - 60 38 net 404 46 63 47 312 201 Read -out Receiver, alignment Receiver, superheterodyne Receiver, transistor Receivers, High -frequency " 109 139 59 m. 180, 233 208 103, 529 526 529 564 555 " " Transceiver, 10 -15 m. Transistorized b.c. 539 529 229 317 Receivers, UHF Reception, Reception, Reception, Recording, Recording, Recording, Recording, frequency modulation mobile 347 Rectification, audio Rectification, stray " Hash " Mercury vapor Selenium Silicon Type meter " " " " " " - polyphase Vacuum Voltage doubler Voltage quadrupler V.t.v.m. Reflected impedance Reflected voltage, transmission line Reflection, ionospheric Ref lectometer, coaxial Reflectometer, SWR meter Regeneration Regulated power supply Regulation, power line Regulation, power supply Regulation, voltage Regulator noise Regulator tube Regulator tube (VR) Regulator, voltage Rejection notch, I -F Rel, def. of Relaxation oscillator Reluctance, def. of Remote cut -off tube Residual magnetism 1. Capacitance filter Dynamic Gain Ground Input Load Plate - - parallel Filter Impedance Mode feedback ground R -F shielding Rhombic antenna Rhumbatron cavity RIAA equalizer curve Ribbon TV line "Ribbon" (TV) transmission line Ring diode modulator Ripple factor, filter Ripple voltage RL circuit R -L integrator network RL transient RLC circuits Root- mean -square (a.c.) Rotary beam antenna Rotator, antenna Ruggedized tube R -F 78 -Q linear amplifier Safety bleeder S- curve, 406 107, 214 series 25 119 73, 74 400 29 23 27, 709 352 527 34 series 353 732 661 24 47 400 56 53, 54 688 54 230 61 2 611, 612 609 607, 608 608 609 604 608 606 270, 357 736 272 358 358 460, 482 230 137 474 418 329 688 686 40 199 38 48 45 490 508 88 S 37 95 94 - chokes dummy loads R -F 36 102, 188 36 Resistor multiplier Resistor, non -inductive, use of Resistor, typical Resonance Antenna Current -F R -F 686 524 686 709 687 220 Radiation Resistive power Resistivity, table of Resistor, bleeder Resistor, characteristics of Resistor, color code Resistor, equalizing Resistor, inductance of Curve R 384 715 - 608 Grounded -grid Grounded screen Inductive tuning Instability neutralization Oscillation Pi- network Push -pull Self -neutralization Tetrode - 1. 687 parallel 166 168 B Construction 690 689 689 694 693, 694 695 696 734 693 695 695 133 57, 63 743 414 742 742 206 23 Internal " Class 71 2 Resistance " R -F 45 Rectifier, bridge " Full -wave " Half -wave " alignment amplifier " Class AB' R -F 379 379 Rectified A.C. 360 140 230 53, 54 420 112, 136 172 234 211, 249 Response, audio Return trace constant amplitude, velocity ....136, 137 136 crossover point 136 high fidelity pre- emphasis 137 RIAA curve - 181 " Oscilloscope pattern Parasitic " Speaker Resonant cavity Resonant circuit Resonant transmission line 511 SSB " Safety precautions Saturation current Saturation, magnetic Sawtooth wave Scalar notation Scale, musical Scatter, ionospheric Scatter signals Screen, CRT Screen grid keying Screen grid tube Screen lead inductance Screen modulation Screen supply Secondary emission Secondary transformer Selective fading, SSB Selectivity, arithmetical Selectivity chart Selectivity control, I -F Selectivity, mobile reception Selectivity, resonant circuit Selectivity, tuned circuits Selenium rectifier Self inductance 805 www.americanradiohistory.com 166 391 389 72 36 59, 172 48 135 414 415 87 395 77 255 286 267 71, 77 61 327 209 338 219 513 54 339 695 37 Self neutralization amplifier Selsyn indicator Semi -conductor Semi -conductor, heat sink Semi- conductor rectifier Semi -conductor, zener Series -cathode modulator Series circuit Series -derived filter Series -diode limiter Series feed Series -feed amplifier Series -parallel circuit resonance Series resistance - Shape factor, -F Shielding, R -F Shot effect Shunt -derived filter Shunt loading, a.v.c. Shunt, meter Shunt neutralization Shunt -regulated bias supply Sidebands, def. of Signal, error Signal, phantom SSB Signal -to- distortion ratio Signal -to -noise ratio Silicon crystal noise generator Silicon rectifier Sine wave Single -ended amplifier Single sideband (S513) " Envelopes " Filter " Jr. exciter " Measurements " Reception " Transmission Single signal reception Single swing oscillator Single -wire antenna tuner Single -wire feed, antenna Single -wire feeder Six- shooter antenna Skeleton VHF antenna Skin effect Skip distance Sky wave Sleeve antenna Slide -wire bridge Slope detection Slotted transmission line S -meter circuits Soldering techniques Sound in air Space charge Space wave Speaker response Speaker tweeter Specific resistance Speech amplifier construction I - Speech Speech Speech Speech Speech clipper, circuitry clipper filter clipping filter, high level waveforms 608 510 90, 21 104 695 105 295 24 64 185 271 604 25 25, 53 217 358 153 64 224 731 252 712 280 193 378 153, 340 152 749 696 42, 43 603 283, 323, 325 326 324 342 150 347 323 220 189, 190 452 437 426 468 - Standard Standard Standard Standing Standing frequency frequency, WWV 208 740 224 727 134 71, 78 410 140 139 23 677 678 302 124, 298, 670 741 525 166 524 Static, wheel Steering diode Step -by -step counter Sterba antenna Storage time, diode Storm, ionospheric Stray capacitance 103 190 465 103 "Strip line" amplifier Stub match, antenna Summation amplifier Summation voltage Summing amplifier Sunspot cycle Superheterodyne receiver Super- regenerative detector Suppression, audio Suppression circuits, parasitics Suppressor Suppressor modulation Suppressor, splatter Surface wave Surge impedance Susceptance, def. of Sweep frequency Switching, transistor action bridge meter, antennascope meter, balanced line meter, bridge meter, reflectometer meter, "twin lamp" Symmetry, amplifier Synchronization, generator Synchronizing voltage Synchronous detection, 55 414 410 476 738 399, 419 Static, corona Static plate current SWR SWR SWR SWR SWR SWR - DSB 806 www.americanradiohistory.com 415 216 595 499 200 197 201 414, 415 208 207 304 362 77 290 300 410 420 52 172 704 454, 746 747 745 741 742 745 605 173 172 349 T capacitance circuit efficiency circuit impedance loading circuit Q Technician class amateur license Teletype, radio Television interference Temperature coefficient Ten -A, SSB exciter Terman- Woodyard amplifier Termination, transmission line Termination, VHF rhombic Test equipment Tetrode, grounded grid use Tetrode modulator Tetrode, zero -bias Tetrode neutralization Tetrode r -f amplifier Tetrode tube Thermal agitation noise Thermionic emission Thermocouple meter Three -phase power supply Threshold voltage Thyratron tube Time -base generator Tank Tank Tank Tank 750 494, 464 739 739 135 pitch wave wave indicator 477 303 283, 292, 299 302, 669 Splatter suppressor 414 Sporadic -E propagation Spotter, frequency 740 Spurious emission spurious pickup 379, 380 Spurious products, mixer 339 Square wave square wave test 58, 61, 62 - - monitor oscilloscope stacked dipole Stacked antenna SSB, - 259 55, 57, 58 260, 262 156 12 322 367 32 343 296 421 483 731 614 669 683 254 606 77 188 67 734, 736 517 696 87 171 Time constant Time sequence keying T -match antenna 38, 60 396 497 limiter Tools, radio T.P.T.G. oscillator 228 TNS - Transceivers, High Frequency " 10 -15 m. " 10 m. mobile Transconductance Transducer pickup Transformation, impedance - Transformation ratio Transformer ampere -turns " Antenna match " Auto " Coupling " " I -F - " Linear matching Matching impedance match " Modulator " Power Transformerless power supply Transient circuit (a.c.) Transient distortion Transient, RC, RL transient wave Transistor - " Bias Code oscillator Complementary circuit Drift Equivalent circuit Heat sink Impedance Junction Mixer Multivibrator Oscillator " " " Point -contact Power Receiver Switching action 6 m, transmitter Transistorized broadcast receiver Transistorized mobile supply Transistorized modulator Transit time Transitron oscillator Transmission line antennascope " Balanced, SWR meter Characteristics Chart Circuits Coaxial Discontinuities Impedance Incident voltage Measurements Reflected voltage Resonant non- resonant Slotted Termination - - " VHF - Kilowatt 721 cathode ray Tracking capacitor Transceiver Trace Transmitter, High- Frequency " Deluxe 200 -watt " 813 grounded -grid amplifier " 350 -Watt grounded -grid Transmitter Control circuits " Design " Ground " Keying " Low Power " Oscilloscope, monitor of 4 -400A "Strip- line" amplifier Transistorized, 6 m. "Tri- band" amplifier 3 -1000Z amplifier 4CX1000A amplifier 4 -400A amplifier Trapezoidal pattern, oscilloscope Trap -type antenna Travelling wave tube Travis discriminator Triode amplifier Triode mixer Triode power amplifier Triode tube Tripler, push -pull Triplex antenna Tropospheric propagation T- section filter Tube, vacuum Tube, VHF design Tubular transmission line Tuned circuit, coaxial Tuned circuit, I -F Tuned circuit, r.f. amplifier Tuned circuit Q Tuned circuits, selectivity Tuner, antenna Tuning, bandspread Tuning indicators Tuning, inductive Turnstile antenna 135 38, 58 90, 92 96 19 97 93 95 104, 707 97 TVI 91 filters TVI -proof enclosures TVI suppression TVI -type filter 100 102 101 - openings "Twin lamp" SWR meter Two -band Marconi antenna 529 704 577 529 581 627 617 649 634 595 577 622 661 654 643 178 510 84 318 110 211 118 72 258 471 411 64 67 232 418 230 216 153 214 339 452 215 224 609 476 369, 374 724, 725 371 Tweeter speaker 95 99 65 139 745 434 u Ultra- linear amplifier Unidirectional antenna Uni -potential cathode Unity coupling Unloaded Q 703 104 81 241 747 745 418 418 229 419 amplifier KW -2 grounded -grid amplifier 239 86, 172 215 528 526 539 555 74, 79 137, 138 63 62 61, 62 498 63 113 209 442 425, 63 293 709 694 59 576 142 500 70 269 57 V Vacuum capacitor Vacuum tube " Amplification 73, 74 Beam power 78 84 107, 87 73 - 421 417 742 tl 740, 744 743 420, 417 740 421 474 Cathode ray Classes classification Conductance Constant- current curve Diode Equivalent noise resistance Foreign Gas 387 is 352 390 391 576 179 356 67 Input loading Klystron Load line Magic eye Magnetron Mixer Operation 807 www.americanradiohistory.com 157 71 153 89 87 152 81 74 88 83 79 75 " " Parameters Pentode Plate resistance Polarity reversal " Remote cutoff " " Vacuum, Shot effect " Space charge " Tetro.Je " " " " " " Thyrtron Travelling wave Triode Upper frequency Variable mu (A) VHF " Voltage regulator " Voltmeter V- Antenna Variable -mu (11) tube Variable reluctance pickup Variac autotronsformer Vector, sine -wave Vehicular noise suppression Velocity modulation Vertical antenna Vertical directivity VHF amplifier " " 106 77 73, 74 76 78 153 89 77 87 84 72 230 154 80 87 130, 734 458 154 137 386 43 523 81 426 407 212 473 475 474 Antennas Antenna polarization " Antenna relay 14 " Bands, characteristics 481 " Corner reflector antenna 473 " Definition of " Discone antenna -ground -plane antenna -.477 " Helical antenna -Horn antenna -- -.479, 482 484 " Multi- element antenna 478 " Nondirectional antenna 362 " Parasitics 474 radiation pattern " Radiation angle 482 " Rhombic antenna 487 " Screen antenna 476 " Sleeve antenna 595 " "Strip -line" amplifier 474 " Transmission line 476 " Turnstile antenna 475 " Wavelength table 698 Vibrator, split -reed 112, 128 Video amplifier 22 Volt, def. of 110 Amplifier Voltage 697 " Avalanche 203 " Break 32 " Breakdown 40 " Decay 26, 52 " Divider 117 " Divider, phase inverter 695 " Doubler supply 28 " Drop, summation 39 Gradient 742 Incident 211 Injection 203 Initial condition 44 Instantaneous 693 Inverse Measurement 731, 733 Multiplier power supply 695 151 Noise 685 Output Quadrupler supply 695 Reflected 742 Regulation 686, 687 Regulator tube 87, 686, 709 Resonant 54 - - " 686 " 197 172 Ripple Summation " Synchronizing " Threshold Voltmeter Voltmeter, A -C V -T Voltmeter, Diode Voltmeter, vacuum tube Volt- ohmmeters Volume compression 696 732 734 734 130, 734 732 670 w Wagner ground Watt, def. of Wave Carrier " Ground - " " " " " " 739 29 205 - -- 410 Harmonic nonsinusoidal Pattern, oscilloscope Peaked sawtooth 58 180 59 Sky space Square transient Surface 410 58 410 59, 175 283, 299, 670 Waveform Waveform, speech Wavelength, def. of Wavelength table, VHF Wave- shaping circuits 401 475 185 24 Weston cell Wheatstone bridge Wheel static Wien- bridge oscillator Williamson amplifier Wire, litz Wiring hints Wiring technique, audio Workshop practice layout W8JK antenna WWV Transmissions - 738 524 191 141 55 527 142 720, 729 469 739 x X -array antenna 465 Y Y, (admittance, def. - of) 52 Yogi, adjustment Yagi antenna Feed systems Yogi antenna, VHF Yogi, constructions Yoke match 505 494 484 502 494 z Zener diode Zeppelin antenna Zero -bias modulator Zero -bias tetrode 808 www.americanradiohistory.com 105 423 683 683 CONVERT SURPLUS RADIO GEAR INTO USEFUL EQUIPMENT -3 volumes give complete conversion data, including instructions, photos and diagrams - - $3.00 ea. 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