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RT8812A

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Dual-Phase PWM Controller with PWM-VID Reference

General Description

The RT8812A is a 2/1 phase synchronous Buck PWM

controller which is optimized for high performance graphic

microprocessor and computer applications. The IC

integrates a Constant-On-Time (COT) PWM controller, two

MOSFET drivers with internal bootstrap diodes, as well

as channel current balance and protection functions

including Over Voltage Protection (OVP), Under Voltage

Protection (UVP), current limit, and thermal shutdown into

the WQFN-20L 3x3 package.

The RT8812A adopts RDS(ON) current sensing technique.

Current limit is accomplished through continuous inductor-

current-sense, while RDS(ON) current sensing is used for

accurate channel current balance. Using the method of

current sampling utilizes the best advantages of each

technique.

The RT8812A features external reference input and PWM-

VID dynamic output voltage control, in which the feedback

voltage is regulated and tracks external input reference

voltage. Other features include adjustable switching

frequency, dynamic phase number control, internal/external

soft-start, power good indicator, and enable functions.

Features

-Dual-Phase PWM Controller

-Two Embedded MOSFET Drivers and Embedded

Switching Boot Diode

-External Reference Input Control

-PWM-VID Dynamic Voltage Control

-Dynamic Phase Number Control

-Lossless RDS(ON) Current Sensing for Current Balance

-Internal Fixed and External Adjustable Soft-Start

-Adjustable Current Limit Threshold

-Adjustable Switching Frequency

-UVP/OVP Protection

-Shoot Through Protection and Short Pulse Free

Technology

-Support an Ultra-Low Output Voltage as Standby

Voltage

-Thermal Shutdown

-Power Good Indicator

-RoHS Compliant and Halogen Free

Simplified Application Circuit

Applications

-CPU/GPU Core Power Supply

-Notebook PC Memory Power Supply

-Chipset/RAM Power Supply

-Generic DC/DC Power Regulator

VOUT

UGATE1

UGATE2

RT8812A

BOOT2

LGATE1

PHASE2

BOOT1

PHASE1

LGATE2

VPVCC

GND RGND VGND_SNS

PVCC

TON

VIN

PGOOD

PSI

VID

VIN

VSNS VOUT_SNS

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Function Pin Description

Pin No. Pin Name Pin Function

1 BOOT1 Bootstrap Supply for PWM 1. This pin powers the high side MOSFET driver.

2 UGATE1

High Side Gate Driver of PWM 1. This pin provides the gate drive for the converter's

high side MOSFET. Connect this pin to the Gate of high side MOSFET.

3 EN Enable Control Input. Active high input.

4 PSI

Power Saving Interface. When the voltage is pulled below 0.8V, the device will operate

into 1 phase DEM. When the voltage is between 1.2V to 1.8V, the device will operate

into 1 phase force CCM. When the voltage is between 2.4V to 5.5V, the device will

operate into 2 phase force CCM.

5 VID Programming Output Voltage Control Input. Refer to PWM-VID Dynamic Voltage

Control.

6 REFADJ Reference Adjustment Output. Refer to PWM-VID Dynamic Voltage Control.

7 REFIN External Reference Input.

8 VREF Reference Voltage Output. This is a high precision voltage reference (2V) from the

VREF pin to RGND pin.

9 TON ON-Time/Switching Frequency Adjustment Input. Connect a 100pF ceramic capacitor

between CTON and ground is optional for noise immunity enhancement.

10 RGND Negative Remote Sense Input. Connect this pin to the ground of output load.

11 SS Soft-Start Time Setting. Connect an external capacitor to adjust soft-start time.

When the external capacitor is removed, the internal soft-start function will be chose.

12 VSNS Positive Remote Sense Input. Connect this pin to the positive terminal of output load.

13 PGOOD Power Good Indicator Output. Active high open drain output.

14 UGATE2

High Side Gate Driver of PWM 2. This pin provides the gate drive for the converter's

high side MOSFET. Connect this pin to the Gate of high side MOSFET.

Marking Information

0Z=YM

DNN

0Z= : Product Code

YMDNN : Date Code

Ordering Information

Note :

Richtek products are :

 RoHS compliant and compatible with the current require-

ments of IPC/JEDEC J-STD-020.

 Suitable for use in SnPb or Pb-free soldering processes.

Pin Configurations

(TOP VIEW)

WQFN-20L 3x3

PSI

BOOT1

UGATE1

BOOT2

UGATE2

VSNS

PGOOD

REFADJ

REFIN

TON

PHASE1

LGATE1

LGATE2

PVCC

17181920

9876

GND

115

PHASE2

VID SS

RGND

VREF

Package Type

QW : WQFN-20L 3x3 (W-Type)

Lead Plating System

G : Green (Halogen Free and Pb Free)

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Function Block Diagram

Pin No. Pin Name Pin Function

15 BOOT2 Bootstrap Supply for PWM 2. This pin powers the high side MOSFET driver.

16 PHASE2

Switch Node for PWM2. This pin is return node of the high side driver of PWM 2.

Connect this pin to the Source of high side MOSFET together with the Drain of

low side MOSFET and the inductor.

17 LGATE2

Low Side Gate Driver of PWM 2. This pin provides the gate drive for the

converter's low side MOSFET. Connect this pin to the Gate of low side MOSFET.

18 PVCC

Supply Voltage Input. Connect this pin to a 5V bias supply. Place a high quality

bypass capacitor from this pin to GND.

19 LGATE1

Low Side Gate Driver of PWM 1. This pin provides the gate drive for the

converter's low side MOSFET. Connect this pin to the Gate of low side MOSFET.

20 PHASE1

Switch Node for PWM1. This pin is return node of the high side driver of PWM 1.

Connect this pin to the Source of high side MOSFET together with the Drain of

low side MOSFET and the inductor.

21 (Exposed Pad) GND Ground. The Exposed pad should be soldered to a large PCB and connected to

GND for maximum thermal dissipation.

LGATE2

PHASE2

UGATE2

BOOT2

LGATE1

PVCC

PHASE1

UGATE1

BOOT1

VSNS

TON

REFIN

VREF

EN Driver

Logic

Boot-Phase

Detection 2

Boot-Phase

Detection 1

S/H

Current

Balance

Control & Protection Logic

TON

Gen 1

TON

Gen 2

PWM1

PWM2

To Power On Reset

VIN

Detection

To Protection Logic

PWM

CMP

To Driver Logic

To Power On Reset

Enable

Logic

40% REFIN

Power On Reset

& Central Logic

Reference

Output Gen.

Soft-Start

& Slew Rate

Control

Threshold

Select

Current

Limit

PGOOD

VID

PSI

REFADJ Mode Select

To Driver LogicZCDPHASE1

RGND

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Operation

The RT8812A is a 2/1 phase synchronous Buck PWM

controller with integrated drivers which are optimized for

high performance graphic microprocessor and computer

applications. The IC integrates a COT (Constant-On-Time)

PWM controller with two MOSFET drivers, as well as

output current monitoring and protection functions.

Referring to the function block diagram of TON Genx, the

synchronous UGATE driver is turned on at the beginning

of each cycle. After the internal one-shot timer expires,

the UGATE driver will be turned off. The pulse width of

this one-shot is determined by the converter's input voltage

and the output voltage to keep the frequency fairly constant

over the input voltage range and output voltage. Another

one-shot sets a minimum off-time.

The RT8812A also features a PWM-VID dynamic voltage

control circuit driven by the pulse width modulation

method. This circuit reduces the device pin count and

enables a wide dynamic voltage range.

Soft-Start (SS)

For internal soft-start function, an internal current source

charges an internal capacitor to build the soft-start ramp

voltage. The output voltage will track the internal ramp

voltage during soft-start interval. For external soft-start

function, an additional capacitor connected from SS pin

to the GND will be charged by a current source and

determines the soft-start time.

PGOOD

The power good output is an open drain architecture.

When the soft-start is finished, the PGOOD open drain

output will be high impedance.

Current Balance

The RT8812A implements internal current balance

mechanism in the current loop. The RT8812A senses per

phase current and compares it with the average current. If

the sensed current of any particular phase is higher than

average current, the on-time of this phase will be adjusted

to be shorter.

Current Limit

The current limit circuit employs a unique “valley” current

sensing algorithm. If the magnitude of the current sense

signal at PHASE is above the current limit threshold, the

PWM is not allowed to initiate a new cycle. Thus, the

current to the load exceeds average output inductor

current, the output voltage falls and eventually crosses

the under voltage protection threshold, inducing IC

shutdown.

Over Voltage Protection (OVP) & Under Voltage

Protection (UVP)

The output voltage is continuously monitored for over

voltage and under voltage protection. When the output

voltage exceeds its set voltage threshold (If VREFIN ≤ 1.33V,

OV = 2V, or VREFIN > 1.33V, OV = 1.5 x VREFIN), UGATE

goes low and LGATE is forced high; when it is less than

40% of its set voltage, under voltage protection is triggered

and then both UGATE and LGATE gate drivers are forced

low. The controller is latched until PVCC is re-supplied

and exceeds the POR rising threshold voltage or EN is

reset.

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Absolute Maximum Ratings (Note 1)

-TON to GND ------------------------------------------------------------------------------------------------------------------ −0.3V to 30V

-RGND to GND --------------------------------------------------------------------------------------------------------------- −0.7V to 0.7V

-BOOTx to PHASEx -------------------------------------------------------------------------------------------------------- −0.3V to 6V

-PHASEx to GND

DC------------------------------------------------------------------------------------------------------------------------------ −0.3V to 30V

<20ns ------------------------------------------------------------------------------------------------------------------------- −8V to 36V

-UGATEx to PHASEx

DC------------------------------------------------------------------------------------------------------------------------------ −0.3V to 6V

<20ns ------------------------------------------------------------------------------------------------------------------------- −5V to 7.5V

-LGATEx to GND

DC------------------------------------------------------------------------------------------------------------------------------ −0.3V to 6V

<20ns ------------------------------------------------------------------------------------------------------------------------- −2.5V to 7.5V

-Other Pins-------------------------------------------------------------------------------------------------------------------- −0.3V to 6V

-Power Dissipation, PD @ TA = 25°C

WQFN-20L 3x3 ------------------------------------------------------------------------------------------------------------- 3.33W

-Package Thermal Resistance (Note 2)

WQFN-20L 3x3, θJA -------------------------------------------------------------------------------------------------------- 30°C/W

WQFN-20L 3x3, θJC ------------------------------------------------------------------------------------------------------- 7.5°C/W

-Lead Temperature (Soldering, 10 sec.)-------------------------------------------------------------------------------- 260°C

-Junction Temperature ------------------------------------------------------------------------------------------------------ 150°C

-Storage Temperature Range --------------------------------------------------------------------------------------------- −65°C to 150°C

-ESD Susceptibility (Note 3)

HBM (Human Body Model) ----------------------------------------------------------------------------------------------- 2kV

Recommended Operating Conditions (Note 4)

-Input Voltage, VIN ----------------------------------------------------------------------------------------------------------- 7V to 26V

-Supply Voltage, VPVCC ---------------------------------------------------------------------------------------------------- 4.5V to 5.5V

-Junction Temperature Range --------------------------------------------------------------------------------------------- −40°C to 125°C

-Ambient Temperature Range --------------------------------------------------------------------------------------------- −40°C to 85°C

Electrical Characteristics

(TA = 25°C unless otherwise specified)

Parameter Symbol Test Conditions Min Typ Max Unit

PWM Controller

PVCC Supply Voltage VPVCC 4.5 -- 5.5 V

PVCC Supply Current ISUPPLY EN = 3.3V, Not Switching -- 1.5 2 mA

PVCC Shutdown Current ISHDN EN = 0V -- -- 10 A

PVCC POR Threshold 3.8 4.1 4.4 V

POR Hysteresis -- 0.3 -- V

Switching Frequency fSW R

TON = 500k (Note 5) 270 300 330 kHz

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Parameter Symbol Test Conditions Min Typ Max Unit

Minimum On-Time tON(MIN) -- 70 -- ns

Minimum Off-Time tOFF(MIN) -- 300 -- ns

EN Input Voltage

Logic-High VENH 1.6 -- --

EN Input Voltage Logic-Low VENL -- -- 0.8

Mode Decision

PSI High Threshold VPSIH Enables Two Phases with FCCM 2.4 -- -- V

PSI Intermediate Threshold VPSIM Enables One Phases with FCCM 1.2 -- 1.8 V

PSI Low Threshold VPSIL Enables One Phases with DEM -- -- 0.8 V

High-Level VVIDH 2 -- --

VID Input Voltage Low-Level VVIDL -- -- 1

Protection Function

Zero Current Crossing

Threshold 8 -- 8 mV

Current Limit Setting Current IOCSET 9 10 11 A

Current Limit Setting Current

Temperature Coefficient IOCSET_TC -- 6300 -- ppm/C

Current Limit Threshold ROCSET = 10k -- 60 -- mV

Absolute Over Voltage

Protection Threshold VOVP, Absolute VREFIN  1.33V 1.9 2 2.1 V

Relative Over Voltage

Protection Threshold VOVP, Relative V

REFIN > 1.33V 145 150 155 %

OV Fault Delay FB forced above OV threshold -- 5 -- s

Relative Under Voltage

Protection Threshold VUVP UVP 35 40 45 %

UV Fault Delay FB forced above UV threshold -- 3 -- s

Thermal Shutdown Threshold TSD -- 150 -- C

PGOOD Blanking Time

(Internal) From EN = high to PGOOD = high

with VSNS within regulation point -- 2.5 -- ms

VSNS Soft-Start (Internal) From first UGATE to VSNS

regulation point, VREFIN = 1V and

VSNS initial = 0V -- 0.7 -- ms

Soft-Start Current Source ISS -- 5 -- A

Error Amplifier

VSNS Error Comparator

Threshold (Valley) V

REFIN = 1V 17.5 12.5 7.5 mV

Reference

Reference Voltage VVREF Sourcing Current = 1mA, VID no

Switching 1.98 2 2.02 V

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Note 1. Stresses beyond those listed “Absolute Maximum Ratings” may cause permanent damage to the device. These are

stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in

the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions may

affect device reliability.

Note 2. θJA is measured at TA = 25°C on a high effective thermal conductivity four-layer test board per JEDEC 51-7. θJC is

measured at the exposed pad of the package.

Note 3. Devices are ESD sensitive. Handling precaution is recommended.

Note 4. The device is not guaranteed to function outside its operating conditions.

Note 5. Not production tested. Test condition is VIN = 8V, VOUT = 1V, IOUT = 20A using application circuit.

Parameter Symbol Test Conditions Min Typ Max Unit

Driver On-Resistance

UGATE Driver Source RUGATEsr BOOTx  PHASEx Forced to 5V -- 2 4 

UGATE Driver Sink RUGATEsk BOOTx  PHASEx Forced to 5V -- 1 2 

LGATE Driver Source RLGATEsr LGATEx, High State -- 1.5 3 

LGATE Driver Sink RLGATEsk LGATEx, Low State -- 0.7 1.5 

From LGATE Falling to UGATE

Rising -- 30 --

Dead-Time

From UGATE Falling to LGATE

Rising -- 20 --

Internal Boost Charging

Switch On-Resistance RBOOT PVCC to BOOTx, IBOOT = 10mA -- 40 80 

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Typical Application Circuit

Figure 1. 2 Active Phase Configuration

Figure 2. 1 Active Phase Configuration

VOUT

UGATE1

UGATE2

RT8812A

REFIN

REFADJ

BOOT2

LGATE1

PHASE2

BOOT1

VREF

PHASE1

LGATE2

VPVCC

VOUT_SNS

GND

21 (Exposed pad)

VSNS

VGND_SNS

18 PVCC

13 PGOOD

PGOOD 4PSI

PSI 5VID

VID

3EN

Enable

VIN

RGND

RSTANDBY

RREF1

RREF2

RREFADJ

RBOOT

CREFIN

CSS

ROCSET

RGND

VSTANDBY

CREFADJ

2.2µF

100k

0.1µF

20k

2k 2.7nF

20k

5.1k 18k NC

0.1µF

10µF x 2

0.36µH/1.05m

10k

47pF

0.1µF

010µF x 2

0.36µH/1.05m 

330µF/2V x 4

22µF x 15

1010

470µF/50V

9TON

VIN

RTON

500k

2.2

1µF CTON

Optional

VOUT

UGATE1

UGATE2

RT8812A

REFIN

REFADJ

BOOT2

LGATE1

PHASE2

BOOT1

VREF

PHASE1

LGATE2

VPVCC

GND

21 (Exposed pad)

18 PVCC

13 PGOOD

PGOOD

4PSI

PSI 5VID

VID

3EN

VIN

RSTANDBY

RREF1

RREF2

RREFADJ

RBOOT

CREFIN

CSS

ROCSET

RGND

VSTANDBY

Floating

VOUT_SNS

VSNS

VGND_SNS

RGND

Enable

CREFADJ

2.2µF

100k

0.1µF

20k

2.7nF

18k NC

5.1k

0.1µF

10µF x 2

0.36µH/1.05m

470µF/50V

22µF x 15

330µF/2V x 4

1010

10k

47pF

9TON

VIN

RTON

500k

2.2

1µF CTON

Optional

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Inductor Current vs. Output Current

0 102030405060

Output Current (A)

Inductor Current (A)

VIN = 19V, VPVCC = 5V

Phase 1

Phase 2

VREF vs. Temperature

1.96

1.97

1.98

1.99

2.00

2.01

2.02

2.03

2.04

-50 -25 0 25 50 75 100 125

Temperature (°C)

VREF (V)

VIN = 19V, VPVCC = 5V, No Load

TON vs. Temperature

165.0

167.5

170.0

172.5

175.0

177.5

180.0

182.5

185.0

-50 -25 0 25 50 75 100 125

Temperature (°C)

TON (ns)

VIN = 19V, VPVCC = 5V, No Load

Typical Operating Characteristics

Efficiency vs. Load Current

10%

20%

30%

40%

50%

60%

70%

80%

90%

100%

0 5 10 15 20 25 30 35 40 45 50 55 60

Load Current (A)

Efficiency (%)

VIN = 19V, VPVCC = 5V,

VOUT = 0.9V, 2 Phase Operation

100

Efficiency vs. Load Current

10%

20%

30%

40%

50%

60%

70%

80%

90%

100%

0.01 0.1 1 10

Load Current (A)

Efficiency (%)

VIN = 19V, VPVCC = 5V,

VOUT = 0.9V, 1 Phase with DEM Operation

100

Power On from EN

Time (1ms/Div)

UGATE1

(50V/Div)

VIN = 19V, VPVCC = 5V, IOUT = 50A

(5V/Div)

UGATE2

(50V/Div)

VOUT

(1V/Div)

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Load Transient Response

Time (20μs/Div)

UGATE1

(50V/Div) VIN = 19V, VPVCC = 5V

IOUT

(50A/Div)

UGATE2

(50V/Div)

VOUT

(100mV/Div)

Dynamic Output Voltage Control

Time (50μs/Div)

UGATE1

(50V/Div)

IOUT = 50A, VREFIN = 1.2V to 0.6V

DVID

(2V/Div)

UGATE2

(50V/Div)

VOUT

(1V/Div)

VIN = 19V, VPVCC = 5V

Power Off from VCC

Time (1ms/Div)

UGATE1

(50V/Div)

VIN = 19V, VPVCC = 5V, IOUT = 50A

PVCC

(5V/Div)

UGATE2

(50V/Div)

VOUT

(1V/Div)

Dynamic Output Voltage Control

Time (50μs/Div)

UGATE1

(50V/Div)

IOUT = 50A, VREFIN = 0.6V to 1.2V

DVID

(2V/Div)

UGATE2

(50V/Div)

VOUT

(1V/Div)

VIN = 19V, VPVCC = 5V

Power On from VCC

Time (1ms/Div)

UGATE1

(50V/Div)

VIN = 19V, VPVCC = 5V, IOUT = 50A

PVCC

(5V/Div)

UGATE2

(50V/Div)

VOUT

(1V/Div)

Power Off from EN

Time (1ms/Div)

UGATE1

(50V/Div)

VIN = 19V, VPVCC = 5V, IOUT = 50A

(5V/Div)

UGATE2

(50V/Div)

VOUT

(1V/Div)

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UVP

Time (20μs/Div)

UGATE1

(20V/Div)

VIN = 19V, VPVCC = 5V, IOUT = 40A

VVSNS

(1V/Div)

LGATE1

(5V/Div)

OCP

Time (20μs/Div)

UGATE1

(50V/Div)

VIN = 19V, VPVCC = 5V

LGATE1

(10V/Div)

IL1

(20A/Div)

IL2

(20A/Div)

OVP

Time (100μs/Div)

UGATE1

(20V/Div)

VIN = 19V, VPVCC = 5V, No Load

VVSNS

(1V/Div)

LGATE1

(5V/Div)

Load Transient Response

Time (20μs/Div)

UGATE1

(50V/Div) VIN = 19V, VPVCC = 5V

IOUT

(50A/Div)

UGATE2

(50V/Div)

VOUT

(100mV/Div)

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Application Information

The RT8812A is a dual-phase synchronous Buck PWM

controller with integrated drivers which is optimized for

high performance graphic microprocessor and computer

applications. A COT (Constant-On-Time) PWM controller

and two MOSFET drivers with internal bootstrap diodes

are integrated so that the external circuit can be easily

designed and the number of component is reduced.

The topology solves the poor load transient response timing

problems of fixed-frequency mode PWM and avoids the

problems caused by widely varying switching frequencies

in conventional constant on-time and constant off-time

PWM schemes.

The IC supports dynamic mode transition function with

various operating states, which include dual-phase with

CCM operation and single phase with diode emulation

mode. These different operating states make the system

efficiency as high as possible.

The RT8812A provides a PWM-VID dynamic control

operation in which the feedback voltage is regulated and

tracks external input reference voltage. It also features

complete fault protection functions including over voltage,

under voltage and current limit.

Remote Sense

The RT8812A uses the remote sense path (VSNS and

RGND) to overcome voltage drops in the power lines by

sensing the voltage directly at the end of GPU. Normally,

to protect remote sense path disconnecting, there are

two resistors (RLocal) connecting between local sense path

and remote sense path. That is, in application with remote

sense, the RLocal is recommended to be 10Ω to 100Ω. If

no need of remote sense, the RLocal is recommended to

be 0Ω.

OUT

ON TON

2 V 3.2p

T = R

V0.5











SOUT INON

F= V V T

And then the switching frequency FS is :

PWM Operation

The RT8812A integrates a Constant-On-Time (COT) PWM

controller, and the controller provides the PWM signal

which relies on the output ripple voltage comparing with

internal reference voltage as shown in Figure 4. Referring

to the function block diagram of TON Genx, the

synchronous UGATE driver is turned on at the beginning

of each cycle. After the internal one-shot timer expires,

the UGATE driver will be turned off. The pulse width of

this one-shot is determined by the converter's input voltage

and the output voltage to keep the frequency fairly constant

over the input voltage and output voltage range. Another

one-shot sets a minimum off-time.

Figure 3. Output Voltage Sensing

RTON is a resistor connected from the VIN to TON pin. The

value of RTON can be selected according to Figure 5.

The recommend operation frequency range is 150kHz to

600kHz.

On-Time Control

The on-time one-shot comparator has two inputs. One

input monitors the output voltage, while the other input

samples the input voltage and converts it to a current.

This input voltage proportional current is used to charge

an internal on-time capacitor. The on-time is the time

required for the voltage on this capacitor to charge from

zero volts to VOUT, thereby making the on-time of the

high side switch directly proportional to output voltage

and inversely proportional to input voltage. The

implementation results in a nearly constant switching

frequency without the need for a clock generator.

VOUT

UGATE

LGATE

BOOT

PHASE

GPU-

VSNS

GPU+

VIN

RGND

RLocal+RLocal-

Local Sense Path

Remote Sense Path

Figure 4. Constant On-Time PWM Control

VPEAK

VOUT

VVALLEY

tON

VREF

VOUT

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Figure 5. Frequency vs. RTON

200

300

400

500

600

700

800

150 250 350 450 550 650 750

RTON (k )

Frequency (kHz) 1

Operation Phase Number PSI Voltage Setting

1 phase with DEM 0V to 0.8V

1 phase with CCM 1.2V to 1.8V

2 phase with CCM 2.4V to 5.5V

Table 1

Active Phase Circuit setting

The RT8812A operates as active phase being 2 phase,

and 1 phase. When programming active phase being 1

phase, The UGATE2, BOOT2, PHASE2, and LGATE 2

pins are floating. As programming active phase being 1

phase, the voltage setting at PSI pin can't higher than

1.8V.

Mode Selection

The RT8812A can operate into 2 phases with force CCM,

1 phase with force CCM, and 1 phase with DEM according

to PSI voltage setting. If PSI voltage is pulled below 0.8V,

the controller will operate into 1 phase with DEM. In DEM

operation, the RT8812A automatically reduces the

operation frequency at light load conditions for saving power

loss. If PSI voltage is pulled between 1.2V to 1.8V, the

controller will switch operation into 1 phase with force

CCM. If PSI voltage is pulled between 2.4V to 5.5V, the

controller will switch operation into 2 phase with force

CCM. The operation mode is summarized in Table 1.

Moreover, the PSI pin is valid after POR of VR.

Diode-Emulation Mode

In diode-emulation mode, the RT8812A automatically

reduces switching frequency at light-load conditions to

maintain high efficiency. As the output current decreases

from heavy-load condition, the inductor current is also

reduced, and eventually comes to the point that its valley

touches zero current, which is the boundary between

continuous conduction and discontinuous conduction

modes. By emulating the behavior of diodes, the low side

MOSFET allows only partial of negative current when the

inductor freewheeling current reaches negative value. As

the load current is further decreased, it takes a longer

time to discharge the output capacitor to the level that

requires the next “ON” cycle. In reverse, when the output

current increases from light load to heavy load, the

switching frequency increases to the preset value as the

inductor current reaches the continuous conduction

condition. The transition load point to the light load

operation is shown in Figure 6 and can be calculated as

follows :

Figure 6. Boundary condition of CCM/DEM

IN OUT

LOAD(SKIP) ON

(V V )





where tON is on-time.

The switching waveforms may be noisy and asynchronous

in light loading diode-emulation operation condition, but

this is a normal operating condition that results in high

light-load efficiency. Trade-off in DEM noise vs. light-load

efficiency is made by varying the inductor value. Generally,

low inductor values produce a broad high efficiency range

vs. load curve, while higher values result in higher full load

efficiency (assuming that the coil resistance remains fixed)

and less output voltage ripple. The disadvantages for using

higher inductor values include larger physical size and

degraded load-transient response (especially at low input

voltage levels).

IPEAK

ILOAD = IPEAK/2

tON

Slope = (VIN - VOUT) / L

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Figure 7. Internal Soft-Start Sequence

The RT8812A also provides an external soft-start function,

and the external soft-start sequence is shown in Figure

8. The external capacitor connected from SS pin to GND

is charged by a 5μA current source to build a soft-start

voltage ramp. If the external soft-start function is chosen,

the external soft-start time should be set longer than

internal soft-start time to avoid output voltage tracking the

internal soft-start ramp. The recommend external soft-start

slew rate is from 0.1V/ms to 0.4V/ms.

Forced-CCM Mode

The low noise, forced-CCM mode disables the zero-

crossing comparator, which controls the low side switch

on-time. This causes the low side gate drive waveform to

be the complement of the high side gate drive waveform.

This in turn causes the inductor current to reverse at light

loads as the PWM loop to maintain a duty ratio VOUT/VIN.

The benefit of forced-CCM mode is to keep the switching

frequency fairly constant.

Enable and Disable

The EN pin is a high impedance input that allows power

sequencing between the controller bias voltage and another

voltage rail. The RT8812A remains in shutdown if the EN

pin is lower than 800mV. When the EN voltage rises above

the 1.6V high level threshold, the RT8812A will begin a

new initialization and soft-start cycle.

Power On Reset (POR), UVLO

Power On Reset (POR) occurs when VPVCC rises above

to approximately 4.1V (typical), the RT8812A will reset

the fault latch circuit and prepare for PWM operation. When

the VPVCC is lower than 3.8V (typical), the Under Voltage

Lockout (UVLO) circuitry inhibits switching by keeping

UGATE and LGATE low.

Soft-Start

The RT8812A provides both internal soft-start function and

external soft-start function. The soft-start function is used

to prevent large inrush current and output voltage overshoot

while the converter is being powered up. The soft-start

function automatically begins once the chip is enabled.

There is a delay time around 1.1ms from EN goes high to

VOUT begins to ramp-up.

If the external capacitor between the SS pin and ground is

removed, the internal soft-start function will be chosen.

An internal current source charges the internal soft-start

capacitor so that the internal soft-start voltage ramps up

linearly The output voltage will track the internal soft-start

voltage during the soft-start interval. After the internal soft-

start voltage exceeds the REFIN voltage, the output voltage

no longer tracks the internal soft-start voltage but follows

the REFIN voltage. Therefore, the duty cycle of the UGATE

signal as well as the input current at power up are limited.

The soft-start process is finished until both the internal

SSOK and external SSOK go high and protection is not

triggered. Figure 7 shows the internal soft-start sequence.

Internal SS

LGATE

UGATE

PGOOD

Current Limit

Programming

VOUT

Soft-Start Normal

External SS

Internal SSOK

External SSOK

PVCC

Soft

Discharged

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Figure 10. PWM VID Analog Circuit Diagram

Figure 11. PWM VID Time Diagram

With the external circuit and VID control signal, the

controller provides three operation modes shown as Figure

11.

VREF

PWM VID

REFIN

BOOT

MODE

NORMAL

MODE BOOT

MODE STANDBY

MODE

STANDBY

CONTROL

Power Good Output (PGOOD)

The PGOOD pin is an open drain output, and it requires a

pull-up resistor. During soft-start, the PGOOD is held low

and is allowed to be pulled high after VOUT achieved over

UVP threshold and under OVP threshold. In additional, if

any protection is triggered during operation, the PGOOD

will be pulled low immediately.

PWM VID and Dynamic Output Voltage Control

The RT8812A features a PWM VID input for dynamic output

voltage control as shown in Figure 10, which reduces the

number of device pin and enables a wide dynamic voltage

range. The output voltage is determined by the applied

voltage on the REFIN pin. The PWM duty cycle determines

the variable output voltage at REFIN.

Figure 8. External Soft-Start Sequence

Figure 9. External Soft-Start Time Setting

The soft-start time can be calculated as :

Where ISS = 5μA (typ.), VREFIN is the voltage of REFIN

pin, and CSS is the external capacitor placed from SS to

GND.

SS REFIN

SS SS

(C V )

t = I



VOUT

tSS

VREFIN

ISS

CSS

VCC

Internal SS

LGATE

UGATE

PGOOD

Current Limit

Programming

Soft-Start Normal

External SS

Internal SSOK

External SSOK

PVCC

VOUT

Soft

Discharged

RGND

REFIN

REFADJ

VREF

VID

PWM IN

CREFIN

RREF2

RBOOT

RREF1 RREFADJ

RSTANDBY

Standby

Mode Control

Buffer

RGND

CREFADJ

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Figure 12. PWM VID Analog Output

Vmin

Vmax

N = 1

N = 2

N = 1

N = 2

N = Nmax

Tvid = Nmax x Tu

VREFIN

VID Duty

00.5 1

VID Input

Boot Mode

VID is not driven, and the buffer output is tri-state. At this

time, turn off the switch Q1 and connect a resistor divider

as shown in Figure 11 that can set the REFIN voltage to

be VBOOT as the following equation :









REF2

BOOT VREF REF1 REF2 BOOT

V = V RRR



STANDBY VREF

REF2 STANDBY

REF1 BOOT REF2 STANDBY

V = V

R// R

R R (R // R )









REF2

min VREF REF2 BOOT

REFADJ BOOT REF2

REF1 REFADJ BOOT REF2

V = V RR

R // (R R )

R R // (R R )



REF2

max VREF REF1 REFADJ BOOT REF2

V = V (R // R ) R R



max min

STEP max

(V V )

V = N

Standby Mode

An external control can provide a very low voltage to meet

VOUT operating in standby mode. If the VID pin is floating

and switch Q1 is enabled as shown in Figure 11, the REFIN

pin can be set for standby voltage according to the

calculation below :

Normal Mode

If the VID pin is driven by a PWM signal and switch Q1 is

disabled as shown in Figure 11, the VREFIN can be adjusted

from Vmin to Vmax, where Vmin is the voltage at zero percent

PWM duty cycle and Vmax is the voltage at one hundred

percent PWM duty cycle. The Vmin and Vmax can be set

by the following equations :

The relationship between VID duty and VREFIN is shown in

Figure 12, and VOUT can be set according to the calculation

below :



OUT min STEP

V = V NV

where VSTEP is the resolution of each voltage step 1 :

where Nmax is the number of total available voltage steps

and N is the number of step at a specific VOUT. The dynamic

voltage VID period (Tvid = Tu x Nmax) is determined by the

unit pulse width (Tu) and the available step number (Nmax).

The recommended Tu is 27ns.

Where VVREF = 2V (typ.)

Choose RREF2 to be approximately 10kΩ, and the RREF1

and RBOOT can be calculated by the following equations :



REF2 VREF BOOT

REF1 BOOT BOOT

REF2 VREF BOOT

REF1 BOOT

BOOT

REF2 VREF BOOT

BOOT REF1

BOOT

RVV

RR V

RVV













By choosing RREF1, RREF2, and RBOOT, the RSTANDBY can

be calculated by the following equation :



STANDBY

REF2 REF1 BOOT STANDBY

REF2 REF STANDBY REF1 REF2 BOOT

REF1

RRR V

RVV RRR



 

   



By choosing RREF1, RREF2, and RBOOT, the RREFADJ can be

calculated by the following equation :

REF1 min

REFADJ max min

RVV





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VID Slew Rate Control

In RT8812A, the VREFIN slew rate is proportional to PWM

VID duty, the rising time and falling time are the same. In

In normal mode, the VREFIN slew rate SR can be estimated

by CREFADJ or CREFIN as the following equation :

When choose CREFADJ :











REFIN_Final REFIN_initial

SR REFIN

SR REF1 REFADJ BOOT REF2

(V V ) 80%

SR = 2.2R C

R = R // R R // R

Current Limit

The RT8812A provides cycle-by-cycle current limit control

by detecting the PHASE voltage drop across the low side

MOSFET when it is turned on. The current limit circuit

employs a unique “valley” current sensing algorithm as

shown in Figure 13. If the magnitude of the current sense

signal at PHASE is above the current limit threshold, the

PWM is not allowed to initiate a new cycle.

In order to provide both good accuracy and a cost effective

solution, the RT8812A supports temperature compensated

MOSFET RDS(ON) sensing.

When choose CREFIN :



REFIN_Final REFIN_initial

SR REFADJ

SR REF1 REFADJ BOOT REF2

(V V ) 80%

SR = 2.2R C

R = (R // R ) // (R +R )



The recommend SR is estimated by CREFADJ.

In an over current condition, the current to the load exceeds

the average output inductor current. Thus, the output

voltage falls and eventually crosses the under voltage

protection threshold, inducing IC shutdown.

Figure 13. “Valley” Current Limit

Current Limit Setting

Current limit threshold can be set by a resistor (ROCSET)

between LGATE1 and GND. Once PVCC exceeds the

POR threshold and chip is enabled, an internal current

source IOCSET flows through ROCSET. The voltage across

ROCSET is stored as the current limit protection threshold

VOCSET. The threshold range of VOCSET is 50mV to 400mV.

After that, the current source is switched off.

ROCSET can be determined using the following equation :







VALLEY LGATEDS(ON)

OCSET OCSET

IR 40mV

R = I

where IVALLEY represents the desired inductor limit current

(valley inductor current) and IOCSET is current limit setting

current which has a temperature coefficient to compensate

the temperature dependency of the RDS(ON).

If ROCSET is not present, there is no current path for IOCSET

to build the current limit threshold. In this situation, the

current limit threshold is internally preset to 400mV.

Negative Current Limit

The RT8812A supports cycle-by-cycle negative current

limiting. The absolute value of negative current limit

threshold is the same with the positive current limit

threshold. If negative inductor current is rising to trigger

negative current limit, the low side MOSFET will be turned

off and the current will flow to input side through the body

diode of the high side MOSFET. At this time, output voltage

tends to rise because this protection limits current to

discharge the output capacitor. In order to prevent shutdown

because of over voltage protection, the low side MOSFET

is turned on again 400ns after it is turned off. If the device

hits the negative over current threshold again before output

voltage is discharged to the target level, the low side

MOSFET is turned off and process repeats. It ensures

maximum allowable discharge capability when output

voltage continues to rise. On the other hand, if the output

is discharged to the target level before negative current

threshold is reached, the low side MOSFET is turned off,

the high side MOSFET is then turned on, and the device

keeps normal operation.

IL,PEAK

ILOAD

IL,VALLEY

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RT8812A employs adaptive dead time control scheme to

ensure safe operation without sacrificing efficiency.

Furthermore, elaborate logic circuit is implemented to

prevent cross conduction. For high output current

applications, two power MOSFETs are usually paralleled

to reduce RDS(ON). The gate driver needs to provide more

current to switch on/off these paralleled MOSFETs. Gate

driver with lower source/sink current capability results in

longer rising/falling time in gate signals and higher

switching loss. The RT8812A embeds high current gate

drivers to obtain high efficiency power conversion.

Inductor Selection

Inductor plays an importance role in step-down converters

because the energy from the input power rail is stored in

it and then released to the load. From the viewpoint of

efficiency, the DC Resistance (DCR) of inductor should

be as small as possible to minimize the copper loss. In

additional, the inductor occupies most of the board space

so the size of it is important. Low profile inductors can

save board space especially when the height is limited.

However, low DCR and low profile inductors are usually

not cost effective.

Additionally, higher inductance results in lower ripple

current, which means the lower power loss. However, the

inductor current rising time increases with inductance value.

This means the transient response will be slower. Therefore,

the inductor design is a trade-off between performance,

size and cost.

In general, inductance is designed to let the ripple current

ranges between 20% to 40% of full load current. The

inductance can be calculated using the following equation :





IN OUT OUT

min SW OUT_rated IN

VV V

L = fkI V

where k is the ratio between inductor ripple current and

rated output current.

Input Capacitor Selection

Voltage rating and current rating are the key parameters

in selecting input capacitor. Generally, input capacitor has

a voltage rating 1.5 times greater than the maximum input

voltage is a conservatively safe design.

Current Balance

The RT8812A implements current balance mechanism in

the current loop. The RT8812A senses per phase current

signal and compares it with the average current. If the

sensed current of any particular phase is higher than the

average current, the on-time of this phase will be

decreased.

The current balance accuracy is major related with on-

resistance of low side MOSFET (RLG,DS(ON)). That is, in

practical application, using lower RLG,DS(ON) will reduce

the current balance accuracy.

Output Over Voltage Protection (OVP)

The output voltage can be continuously monitored for over

voltage protection. If REFIN voltage is lower than 1.33V,

the over voltage threshold follows to absolute over voltage

2V. If REFIN voltage is higher than 1.33V, the over voltage

threshold follows relative over voltage 1.5 x VREFIN. When

OVP is triggered, UGATE goes low and LGATE is forced

high. The RT8812A is latched once OVP is triggered and

can only be released by PVCC or EN power on reset. A

5μs delay is used in OVP detection circuit to prevent false

trigger.

Output Under Voltage Protection (UVP)

The output voltage can be continuously monitored for under

voltage protection. When the output voltage is less than

40% of its set voltage, under voltage protection is triggered

and then all UGATE and LGATE gate drivers are forced

low. There is a 3μs delay built in the UVP circuit to prevent

false transitions. During soft-start, the UVP blanking time

is equal to PGOOD blanking time.

MOSFET Gate Driver

The RT8812A integrates high current gate drivers for the

MOSFETs to obtain high efficiency power conversion in

synchronous Buck topology. A dead-time is used to prevent

the crossover conduction for high side and low side

MOSFETs. Because both the two gate signals are off

during the dead-time, the inductor current freewheels

through the body diode of the low side MOSFET. The

freewheeling current and the forward voltage of the body

diode contribute power losses to the converter. The

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The next step is to select proper capacitor for RMS current

rating. Use more than one capacitor with low Equivalent

Series Resistance (ESR) in parallel to form a capacitor

bank is a good design. Besides, placing ceramic capacitor

close to the drain of the high side MOSFET is helpful in

reducing the input voltage ripple at heavy load.

Output Capacitor Selection

The output filter capacitor must have ESR low enough to

meet output ripple and load transient requirement, yet have

high enough ESR to satisfy stability requirements. Also,

the capacitance must be high enough to absorb the inductor

energy going from a full load to no load condition without

tripping the OVP circuit. Organic semiconductor

capacitor(s) or special polymer capacitor(s) are

recommended.

MOSFET Selection

The majority of power loss in the step-down power

conversion is due to the loss in the power MOSFETs. For

low voltage high current applications, the duty cycle of

the high side MOSFET is small. Therefore, the switching

loss of the high side MOSFET is of concern. Power

MOSFETs with lower total gate charge are preferred in

such kind of application.

However, the small duty cycle means the low side MOSFET

is on for most of the switching cycle. Therefore, the

conduction loss tends to dominate the total power loss of

the converter. To improve the overall efficiency, the

MOSFETs with low RDS(ON) are preferred in the circuit

design. In some cases, more than one MOSFET are

connected in parallel to further decrease the on-state

resistance. However, this depends on the low side

MOSFET driver capability and the budget.

OUT OUT

RMS OUT IN IN

I = I 1









The input capacitor is used to supply the input RMS

current, which can be approximately calculated using the

following equation :

Thermal Considerations

For continuous operation, do not exceed absolute

maximum junction temperature. The maximum power

dissipation depends on the thermal resistance of the IC

package, PCB layout, rate of surrounding airflow, and

difference between junction and ambient temperature. The

maximum power dissipation can be calculated by the

following formula :

PD(MAX) = (TJ(MAX) − TA) / θJA

where TJ(MAX) is the maximum junction temperature, TA is

the ambient temperature, and θJA is the junction to ambient

thermal resistance.

For recommended operating condition specifications, the

maximum junction temperature is 125°C. The junction to

ambient thermal resistance, θJA, is layout dependent. For

WQFN-20L 3x3 package, the thermal resistance, θJA, is

30°C/W on a standard JEDEC 51-7 four-layer thermal test

board. The maximum power dissipation at TA = 25°C can

be calculated by the following formula :

PD(MAX) = (125°C − 25°C) / (30°C/W) = 3.33W for

WQFN-20L 3x3 package

The maximum power dissipation depends on the operating

ambient temperature for fixed TJ(MAX) and thermal

resistance, θJA. The derating curve in Figure 14 allows

the designer to see the effect of rising ambient temperature

on the maximum power dissipation.

Figure 14. Derating Curve of Maximum Power

Dissipation

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

0 25 50 75 100 125

Ambient Temperature (°C)

Maximum Power Dissipation (W) 1

Four-Layer PCB

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Layout Considerations

Layout is very important in high frequency switching

converter design. If designed improperly, the PCB could

radiate excessive noise and contribute to the converter

instability. Following layout guidelines must be considered

before starting a layout for RT8812A.

Place the RC filter as close as possible to the PVCC

pin.

Keep current limit setting network as close as possible

to the IC. Routing of the network should avoid coupling

to high voltage switching node.

Connections from the drivers to the respective gate of

the high side or the low side MOSFET should be as

short as possible to reduce stray inductance.

All sensitive analog traces and components such as

VSNS, RGND, EN, PSI, VID, PGOOD, VREF, TON

VREFADJ, VREFIN and TSNS should be placed away

from high voltage switching nodes such as PHASE,

LGATE, UGATE, or BOOT nodes to avoid coupling. Use

internal layer(s) as ground plane(s) and shield the

feedback trace from power traces and components.

Power sections should connect directly to ground

plane(s) using multiple vias as required for current

handling (including the chip power ground connections).

Power components should be placed to minimize loops

and reduce losses.

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Richtek Technology Corporation

14F, No. 8, Tai Yuen 1st Street, Chupei City

Hsinchu, Taiwan, R.O.C.

Tel: (8863)5526789

Richtek products are sold by description only. Richtek reserves the right to change the circuitry and/or specifications without notice at any time. Customers should

obtain the latest relevant information and data sheets before placing orders and should verify that such information is current and complete. Richtek cannot

assume responsibility for use of any circuitry other than circuitry entirely embodied in a Richtek product. Information furnished by Richtek is believed to be

accurate and reliable. However, no responsibility is assumed by Richtek or its subsidiaries for its use; nor for any infringements of patents or other rights of third

parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Richtek or its subsidiaries.

Outline Dimension

Dimensions In Millimeters Dimensions In Inches

Symbol Min Max Min Max

A 0.700 0.800 0.028 0.031

A1 0.000 0.050 0.000 0.002

A3 0.175 0.250 0.007 0.010

b 0.150 0.250 0.006 0.010

D 2.900 3.100 0.114 0.122

D2 1.650 1.750 0.065 0.069

E 2.900 3.100 0.114 0.122

E2 1.650 1.750 0.065 0.069

e 0.400 0.016

L 0.350 0.450

0.014 0.018

W-Type 20L QFN 3x3 Package

Note : The configuration of the Pin #1 identifier is optional,

but must be located within the zone indicated.

DETAIL A

Pin #1 ID and Tie Bar Mark Options

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