1982_Fairchild_High_Current_Voltage_Regulators 1982 Fairchild High Current Voltage Regulators
User Manual: 1982_Fairchild_High_Current_Voltage_Regulators
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I=AIRCHILD A Schlumberger Company 369 Whisman Road, Mountain View, California 94043 ©1982 Fairchild Camera and Instrument Corporation/369 Whisman Road, Mountain View, California 94043/(415) 962-5011/TWX 91 0-379~6435 Fairchild reserves the right to make changes in the circuitry or specifications in this book at any time without notice. Manufactured under one of the following U.S. Patents: 2981877, 3015048, 3064167, 3108359, 3117260, other patents pending. Fairchild reserves the right to make changes in the circuitry or specifications in this book at any time without notice. Manufactured under one of the following U.S. Patents: 2981877, 3015048, 3064167, 3108359, 3117260, other patents pending. Fairchild cannot assume responsibility for use of any circuitry described other than circuitry entirely embodied in a Fairchild product. No other circuit patent licenses are implied. Printed in U.S.A. 610000 100M January 1982 FAIRCHILD Introduction A Schlumberger Company The Fairchild Hybrid Division was established to fulfill the high-volume, high-quality, low-cost requirements of a growing number of companies turning to hybrid technology. A broad range of precision hybrid voltage regulators is available off-the-shelf as well as automotive ignition systems. For custom hybrid programs we offer full design capability with rapid prototyping and translation to volume production. No other hybrid manufacturer in the world can match Fairchild's total capabilities. Since Fairchild is one of the world's leading semiconductor manufacturers, there is no need to depend on outside suppliers for delivery, reliability or quality of semiconductor components. Our years of experience in developing and qualifying sources of passive components for the automotive hybrid market assure dependable performance in the finished product. Facilities in Northern California and Hong Kong can turn out more reliable hybrid products in a day than most hybrid suppliers can produce in a week. Fairchild is equipped to produce hybrids for any company in any business that uses hybrid products in large quantities. The markets served include automotive, consumer electronics, computer, telecommunications, industrial controls, aerospace and military. FAIRCHILD Table of Contents A Schlumberger Company Chapter One Page Capabilities Information Design .................................... 1-3 Production ..................................1-3 Chapter Two Reliability What is Reliability? .......................... 2-3 Some Reliabili~v Terms ....................... 2-3 How is Reliabilicy Obtained? ................... 2-4 Design Piece Parts Wafer Fabrication Assembly Environmental Stresses How is Reliability Tested and Maintained? ........ 2-7 Die Related Tests Package Related Tests Conclusion ................................ 2-8 Chapter Three Cross Reference Guide Ordering Information ......................... 3-3 Chapter Four Data Sheets ILA78H05.ILA78H05A ........................ 4-3 ILA78H12A ................................. 4-7 ILA78HGA ................................ 4-11 ILA78P05 ................................. 4-15 ILA79HG .................................. 4-19 SH323. SH223. SH123 ...................... 4-23 SH1605 .................................. 4-27 Chapter Five Applications High Current Voltage Regulator Applications. . . .. 5-3 Understanding the Switching Regulator ......... 5-16 Power Supply Design ....................... 5-31 Thermal Considerations ..................... 5-41 Chapter Six Fairchild Field Sales Offices Representatives and Distributors .............. 6-3 FAIRCHILD A Schlumberger Company __ l~c ap_a_b_i_lit_ie_S__ln_fo_r_m_a_t_io_n_____________1IIIII 1-2 Chapter 1 Capabilities FAIRCHILD A Schlumberger Company Design Hybrids offer all the advantages of other semiconductor components-small size, high reliability and low cost in high volume-while providing fully tested functions that readily interface with the user's overall system. In addition they offer unique advantages such as ability to mix technologies not achievable monolithically, ability to operate in extremely hostile environments, and improved ratios of reliability to. complexity, while retaining their size advantage over the discrete approach. A broad range of process techniques has been developed from which to select the manufacturing method best suited to a specific objective. This variety of techniques provides for wide design flexibility with minimal process constraints. Also, since the semiconductors used in Fairchild hybrids are almost exclusively supplied internally, parameters can be selected and stringently controlled for compatibility with the overall circuit design. New process development and materials research takes place on a continuous basis to ensure that our methods and materials are selected from the best alternatives available. InternallC design compatibility ~xists for linear and other bipolar technologies, power transistors are also designed within the division. Design and production of custom products follow a thorough routine. Once a user's input is submitted, whether in the form of functional specifications, a circuit diagram or breadboard, a comprehensive cost analysis and circuit evaluation is made. Only when Fairchild is satisfied with the circuit/cost analysis is the paper design submitted. Upon approval, a breadboard is produced for user ~valuation. A complete detailed layout is then constructed for use in producing preliminary parts for customer ~pproval. When the customer gives the go-ahe~d, volume production begins. In either instance, standard or custom, a Fairchild hybrid subsystem, fabricated using mixed technologies, is fully tested and delivered on time, in volume quantities. All materials are pre-tested and qualified before admission to the thick film process; pastes are subjected, on a lot basis, to stringent incoming tests using computer controlled equipment. The flatness and surface finish of the alumina substrates are rigidly controlled. Printing onto these substrates is accomplished using automatic magazine fed machines having a high degree of stability and print alignment. The substrates are then transferred by automatic collation equipment to a drying and firing furnace, in which the environment is moisture and oilfree to ensure the control required for maintenance of tight resistor temperature coefficient of resistance (TCR) distributions. Furnace zones are microprocessor controlled, and interfaced to a computer system capable of providing check profiles on demand. Post-firing offload is accomplished using automatic equipment eliminating handling damage. Base conductor and dielectric layers are individually printed and fired; subsequent resistor prints are dried between applications and finally co-fired to the desired pre-trim values. 'Process monitoring, again using computer controlled equipment, constantly verifies visual, electrical and dry-print thickness measurements, assuring high-yield low-cost production. Tailoring the substrate resistors to final value is achieved using active and passive laser trimming systems with carousel feeds and closedcircuit TV monitoring systems. Extensive computer control atthis step provides flexibility and accuracy. Production Fairchild excels in high volume production of hybrid devices. Facilities include: complete thick film production; rubylith, photo reduction and screen manufacturing; active and passive laser trim; wafer sort and scribe; assembly and packaging; testing and quality control. 1-3 Capabilities Total thick-film production capability encompasses gold and palladium-silver conductor systems, resistor prints in the range 1Q to 10MQ, high quality pinhole-free dielectric systems and multilayer techniques. Fairchild's thick film capability is complemented by wafer sort and scribing facilities employing diamond and saw techniques. Wafers are supplied to this area from the company's integrated circuit and discrete fabrication areas. Linear devices and power transistors can be supplied from the division's internal capability. The proximity of all facilities ensures rapid resolution of any technical or scheduling problems. bondil1g techniques is currently under development. Because of its potential for computer controlled assembly when combined with automatic pick and place equipment, tape carrier appears to be the best solution for high volume production of reliable hybrids at minimum cost. Many other advanced techniques in hybrid technology have been developed over the past decade to meet a variety of customer objectives. This trend will continue. Fairchild is committed to producing the highest quality hybrids possible to endure the most stringent environmental conditions in both commercial and military applications. Hybrid production utilizes all the standard manufacturing methods plus a number of proprietary processes designed to meet exacting customer requirements. For example, an exclusive flip-chip solder reflow process has been developed to eliminate bonding steps in large-volume custom applications, while simultaneously providing an extemely rugged micro-interconnect capable of withstanding wide temperature excursions and the most demanding corrosion and vibration environments. For applications involving the use of LSI chips with large area and I/O counts, a versatile interconnect scheme using the latest tape automated 1-4 FAIRCHILD A Schlumberger Company 2-2 Chapter 2 Reliability FAIRCHILD A Schlumberger Company What Is Reliability Wear-Out Failure Reliability is defined as the behavior of a component, a machine, or a system, as a function of time. Statistically, it is also expressed as the probability that the item will perform a required function under established conditions for a given period of time. This failure occurs as a result of degradation, physical or chemical. This Time is a variable covering minutes, hours, months, or years. Some of the equipment in oil well logging operations must have, on the average, a useful life of only five minutes:At the other extreme, the telephone companies want their equipment to last a minimum of 25 years. MiSSiles are subjected to thousands of "g" (gravity) forces and the components must not lose their monitoring, telemetering, or sensing capabilities during the first critical seconds of their flight. Power supplies are constantly being thermally cycled as they are turned on and off. Voltage regulators undergo similar stresses, and both must provide thousands of hours of flawless operation. • Repeated thermal cycling will cause electrical discontinuities between conductors having different coefficients of thermal expansion. • Voltage incursions may cause shorting across a dielectric. • Excessive current densities may introduce metal migration, causing shorts. • Continuous vibrations may cause loss of contact or create loose conducting particles and subsequent shorting. • Moisture is known to degrade components because of chemical reactions resulting in parameter changes. Figure 1 suggests that, to obtain the lowest failure rate a removal of the infant mortality weaknesses is required. Traditionally, this removal was done, as a rule, on military products and for certain nonmilitary users requiring the highest reliability. Some Reliability Terms Initial/Infant Failure A failure occurring during the early stages of operation, the failure rate during infancy is higher than during long term operation. More recently, the growing complexities of various electronic systems, of multi-million dollar computers, some of which contain as many as a quarter million integrated circuits, have created demands by the commercial users for a reliability level similar to that of the military, the difference being only in the temperature range of operation. Infant failures are caused by weaknesses not removed by the numerous inspection operations, if detectable at all. Random Failure A failure which occurs sometimes between the infant mortality and the wear-out periods; the failure rate during this period is generally constant. Fig. 1 A Diagrammatic Representation of Failure Patterns _100 Eo a:~ 10 III a: :;) ::! I:f 1 i-.-+l-_-----CONSTANT------*_-+l INFANT MORTALITY 2-3 Reliability In Table 1 the major component parts are listed with tests normally performed for conformance, and, for reliability. How Is Reliability Obtained? Design High Current voltage regulators are comprised of three major parts: Active components, substrate and package. Table 1 Parts A linear integrated circuit (L1C) and a power transistor provide all the drives and controls: A start circuit, voltage regulation, current sourcing, shortcircuit protection, thermal shutdown and . amplification. The high current densities typical of high power devices and especially of power transistors, integrated on a chip or discrete, require unif.orm current distribution, especially along the emitter contacts, to prevent current hogging, hot spots and excessive heating. This joule heating can be sufficient to reach the aluminum-silicon eutectic temperature, melt the silicon and short the emitter and collector. This effect is minimized by large geometries that decrease the current densities and by diffusion and concentration profiles tha~ insure better current distribution. Long term aluminum migration, a concern wherever large current densities exist, is also eliminated by the proper "sizing" of the conductio~ lines. -:his concept <;>f safe margins and of conservative design rules applies also to all components procured from outSide vendors. Test and Control of Major Component Piece Part Test/Control Substrate Mechanical dimensions Mechanical strength Thermal conduction properties Electrical insulation Chemical composition Thermal cycling Package Mechanical dimensions and characteristics Solderability of leads Electrical insulation Lead seal check Lead strength Solder Preform Mechanical dimensions Chemical composition Die Visual inspection Capacitor Mechanical dimenSions Electrical properties Electrical stress on a sample Process Control of Wafer Fabrication Operations Wafer fabrication is a very complex and disciplined operation and listing all the numerous control points and monitoring operations is beyond the scope of this Reliability section. Reliability, quality and yield are major concerns for any wafer fab operation, and by exte~sion, o! any manufacturing operation. Only a few Items Will be mentioned here: Materials The substrate provides a base for the thick film reSistors, the connections for attaching the active components, and a means of heat dissipation. • • Wafer purity Analysis of chemicals • • Dust particle concentrations Temperature/humidity • Analysis of dopant sources • Gas flows into furnaces • Furnace profiles • CN plots for checking ionic drift The package connects to the equipment for both testing and usage and gives an .additional path for heat dissipation. 2-4 • Equipment calibration • • Exposure Development • • Etching Cleaning, etc .... Reliability the flow referenced in MIL-STD 883 (Test Methods & Procedures for Microelectronics), Method 5008 (Test Procedures for Hybrids and Multichip Microcircuits). This "hi-rei" flow performs the following: • Storage. Isolates product not capable, for mechanical reasons, of storage at 150°C for 24 hours. • Temperature cycle. Eliminates product exhibiting mechanical damages that would cause functional failures. • Constant acceleration. Eliminates structural and mechanical weaknesses: - Poor wire-to-die bonding Poor substrate-to-package attach Poor die-to-substrate attach • Seal. Prevents the components, active and passive, from being influenced by outside factors of the working environment, mostly humidity. • Burn-In. Screens or eliminates all marginal devices, those with inherent defects resulting from manufacturing, aberrations which cause time and stress dependent failures. In the absence of burn-in, these defective units would result in infant/early mortality failures (see paragraph on "Some Reliability Terms"). A typical assembly flow, shown in Table 2, shows both the operation and its control equivalent. • Environmental Testing An additional level of reliability can be obtained by performing environmental and electrical tests along Electrical testing at temperature extremes. Removes all units not meeting functional and parametric criteria. Standard and hi-rei flows are compared on Table 3. Controi of Assembly Operations Completed wafers are electrically probed, and good dice are identified for assembly. Every assembly operation is critical, and every effort is made to guarantee long term life of the product. Table 2 A Typical Assembly Operation Operation Control Scribing (or sawing) Maintenance Separation of good & bad die Visual inspection, QC sample, conformance inspection Die attach Functional check for adherence and wetting; 100% X-Ray; 1OO%visual check Push Test Substrate attach Functional check for adherence and wetting; 100% X-Ray; 100% visual check Push Test Wire bonding Incoming wiretest Pull strength Visual check Optical check (preseal) QC sample conformance inspection Package seal Hermeticity check for fine & gross leak 100% Optical check (post-seal) QC sample conformance inspection 2-5 Reliability Table 3 A Comparison of Hi Rei and Standard Flows Fairchild Unique Level B Hi Rei Standard o Package Seal Package Seal o Post Seal Visual Inspection Post Seal Visual Inspection Post Seal Sample Inspection Q Post Seal Sample Inspection o Bake - o Temperature Cycling -65°C to 150°C - 10X o Constant Acceleration 10KG - Y1 Axis o Seal- Fine Gross 150°C/24 hrs Seal- Fine Gross Q o o Electrical Test o Electrical Test - Q • 25°C • -55°C • 125°C Quality Conformance Electrical Test Burn In - TJ = 150°C Max Time = 160 Hrs. Min. Post Burn In 1. Electrical • 25°C • -55°C • 125°C 2. Visual/Mechanical Quality Conformance 1. Electrical • 25°C • O°C ·100°C 2. Visual/Mechanical 3. Group B - Re: Mil Std 883 As Applicable 4. Group C - Re: Mil Std 883 As Applicable o = 100% Operation Q = Quality Conformance Inspection 2-6 Reliability How Is Reliability Tested And Maintained? No product will be put on the market unless it meets the stringent reliability requirements determi.!1ed by the procurement agency or by the factory. These requirements can be in terms of FITs (Failures In Time) or percent per thousand hours, quantities that give a mathematical limit to the failure rates resulting from a given stress. They can also be in terms of time, e.g., time to 10%,20%, or 50% failure of a given sample for a given test. MIL-STD 883, previously mentioned, lists both the tests and the frequency of these tests performed to maintain qualification-or suitability for sale-of a given product. New products, new processes, new design, new materials, are all "qualified" along the lines originally established by MIL-STD-883. Two major series of tests designed for periodic monitoring or for original qualification and are listed in Table 4, together with a brief description and the respective LTPD (Lot Tolerance Percent Defective) that gives sample size and allowed failures. Table 4 Qualification and Monitor Testing A. Die Related Tests - Group C Test Description LTPD Temperature Cycling Constant Acceleration Seal Fine Seal Gross Electrical Test -65°C to 150°C 10 kg along Y1 axis Helium or Krypton Fluorocarbon/bubble 15 Operating Life Static and/or dynamic 10 B. Package Related Tests - Group D • Lead Integrity Seal Fine Seal Gross Lead bending See Group C above See Group C above 15 • Thermal Shock Temperature Cycling Moisture Resistance Seal Fine Seal Gross Visual Inspection -55°Ct0125°C 15X -65°Ct0150°C 100X Variable temp/humidity lOx See Group C above See Group C above 15 • Mechanical Shock Constant Acceleration Seal Fine Seal Gross Visual Examination 3000g .3 ms 1kg Y1 axi.s See Group C above See Group C above 15 • Salt Atmosphere Seal Fine Seal Gross Visual Examination Salt Atmosphere at 35°C See Group C above See Group C above 15 2-7 Reliability Conclusion The most critical area in any electrical/electronic system is the power supply. ,If the power supply fails, the system goes down. Power supply failure may result in loss of critical data or damage to other system components. To the equipment user, this means idle labor hours and unexpected replacement, repair and service costs. To the equipment manufacturer, it can mean customer dissatisfaction and excessive warranty and rework costs. The power supply is critical to any system and the heart of the power supply is the voltage regulator. The Fairchild Hybrid Division recognizes the importance of quality and reliability to our customer ... and to his customer. Quality and reliability standards are established, before the product is designed and are rigidly adhered to throughout the production flow. High Current Voltage Regulators are presently shipped to a guaranteed AQL of 0.1%, with an actual return rate far less. Fairchild shares its customers' concern for quality and reliability and will continue to improve its products to insure their equipment achieves optimum performance. 2-8 FAIRCHIL.D A Schlumberger Company 3-2 Chapter 3 Cross Reference Guide and Ordering Information FAIRCHILD A Schlumberger Company Cross Reference Guide Cross Reference Voltage Regulator FixedPositive 0 AdjustablePositive 0 Output Capability Fairchild Device Lambda National Silicon General 5V, 3 A 5V, 5 A 5V, 5 A 5 V, 8 A 5 V, 10 A 12 V, 5 A SH323 78H05 78H05A 78P05 78P05 78H12A LAS1405 LAS1405, 1905 LAS1405,1905 LAS3905 Not Available LAS1412,1912 LM323 Not Available Not Available Not Available Not Available Not Available SG323 Not Available Not Available Not Available Not Available Not Available 5 To 24 V, 3 A 5 To 24 V, 5 A 78HGA 78HGA LAS14U LAS19U LM350 LM338 SG350 Not Avaiiable AdjustableNegative -2 To -24 V, 5 A79HG LAS18U Not Available Not Available AdjustableSwitching 3 To 30 V, 5 A Not Available LH1605 Not Available SH1605 (Step Down) 3-3 Cross Reference Guide Ordering Information Unique Level B Processing. To meet the need for improved reliability in the military market, high current voltage regulators are available with special processing. Devices ordered to this program are subject to 100% screening as outlined in chapter 2. Devices may be ordered by simply adding the letters "08" to the end of the ordering code. Ordering Information Fairchild High Current Voltage Regulators may be ordered using a simplified purchasing code. S C- Temperature Range Code ~ Package Code Device Type (5 to 8 Digits) Example (a) 79 HG SM 08 This number code indicates a 5 amp, adjustable negative voltage regulator, packaged in a steel, 4-lead TO -3 with an operating junction . temperature range of -55°C TO +150°C and screened to the Fairchild unique level 8 program as outlined in Chapter 2. Temperature Range Code Operating Junction Temperature C = Commercial O°C to + 150°C (unless otherwise specified) V = Industrial (SH 223 only) -25°C to +150°C M = Military -55°C to +150°C Package Code S = Steel TO - 3 Package 2-Lead } 4-Lead Refer To Chapter 4 8-Lead Device Type (5 to 8 Digits) SH323 3 A, 5 V Fixed Regulator 78H05A 5 A, 5 V Fixed Regulator SH1605' 5 A Switching Regulator Examples (a) SH 323 SC This number code indicates a 3 amp, 5 volt fixed regulator packaged in a steel, 2-lead TO -3 with an operating junction temperature range of -25°C TO +150°C (b) 78 HG ASM This number code indicates a 5 amp, adjustable regulator with guaranteed maximum dropout voltage limits, packaged in a steel, 4-lead TO -3 with an operating junction temperature range of -55°C to +150°C. 1 . ! 3·4 FAIRCHILD A Schlumberger Company 1IIII ~ID_a_m__S_h_e_et_s____________________ 4-2 JlA78HOS • JlA78H05A FAIRCHILD 5-Volt 5-Amp Voltage Regulators A Schlumberger Company Hybrid Products Connection Diagram TO-3 Metal Package Description The jtA78H05 and jtA78H05A are hybrid regulators with 5.0 V fixed outputs and 5.0 A output capabilities. They have the inherent characteristics of the monolithic 3-terminal regulators, i.e., full thermal overload, short-circuit and safe-area protection. All devices are packaged in hermetically sealed TO-3s providing 50 W power dissipation. If the safe operating area is exceeded, the device shuts down rather than failing or damaging other system components (Note 1). This feature eliminates costly output circuitry and overly conservative heat sinks typical of highcurrent regulators built from discrete components. • • 5.0 A OUTPUT CURRENT INTERNAL CURRENT AND THERMAL OVERLOAD PROTECTION • INTERNAL SHORT CIRCUIT PROTECTION • LOW DROPOUT VOLTAGE (TYPICALLY 2.3 V @ 5.0 A) . • 50 W POWER DISSIPATION • STEEL TO-3 PACKAGE • ALL PIN-FOR-PIN COMPATIBLE WITH THE SH323 (Top View) Order Information Type Package J.lA7805 Metal jtA7805A Metal jtA7805 Metal jtA7805A Metal Note 1. These voltage regulators offer output transistor safe·area protection. However, to maintain full protection. the devices must be operated within the maximum input·to-output voltage differential ratings, as listed on this data sheet under "Absolute Maximum Ratings." For applications violating these limits, devices will not be fully protected. Code GN GN GN Part No. jtA78H05SC jtA78H05ASC jtA78H05SM jtA78H05ASM GN Block Diagram T T I START CIRCUIT CURRENT SOURCE VOLTAGE REGULATOR I -::L. 7 r-- VIN THERMAL SHUTDOWN 1 I SHORT CIRCUIT PROTECTION R "- "- ,..... rK Rsc IJ 2 VOUT 3 COMMON 4-3 JlA78H05 • JlA78H05A Absolute Maximum Ratings Input Voltage Input-to-Output Voltage Differential, Output Short Circuited Internal Power Dissipation Operating Junction Temperature Military Temperature Range ILA78H05SM ILA78H05ASM ILA78H05 • ILA78H05A Electrical Characteristics Commercial Temperature Range ILA78H05SC ILA78H05ASC Storage Temperature Range Pin Temperature (Soldering, 60 s) 40V 35 V 50 W @ 25°C Case 150°C O°Cto +150°C O°C to +150°C -55°C to +150°C 300°C -55°C to +150°C -55°C to +150°C TJ = 25°C, VIN = 10 V, lOUT = 2.0 A unless otherwise specified. Limits Symbol Characteristic Condition Min Typ Max VOUT Output Voltage lOUT = 2.0 A VIN = 8.5 to 25 V (ILA78H05) VIN = 7.5 to 25 V (ILA78H05A) 4.85 5.0 5.25 V 10 50 mV 10 50 mV mV dVOUT Line Regulation (Note 2) dVOUT Load Regulation (Note 2) 10 mA ::5 lOUT ::5 5.0 A 10 50 10 Quiescent Current 3.0 10 RR Ripple Rejection =0 lOUT = 1.0 A, f = 120 Hz, 5.0 Vpk-pk Vn Output Noise 10 Hz ::5 f ::5 100 kHz lOUT dB 40 = 5.0 A lOUT = 3.0 A lOUT = 5.0 A lOUT = 3.0 A Dropout Voltage (Note 3) VOO ILA78H05A Short-Circuit Current Limit lOS rnA 60 lOUT ILA78H05 Unit ILVRMS 2.3 V 2.0 V 2.3 2.5 V 2.0 2.3 V 7.0 12.0 Apk Notes 2. Load and line regulation are specified at constant junction temperature. Pulse testing is required with a pulse width ~ 1 ms and a duty cycle of ~ 5"10. Full Kelvin connection methods must be used to measure these parameters. 3. Dropout Voltage is the input-output voltage differential that causes the output voltage to decrease by 5"10 of its initial value. Typical Performance Curves Output Impedance Output Noise Voltage 100 . Maximum Power Dissipation 80 0, lOUT = 1 A YIN = lOY YIN - 10V CL = 0 50 0.5 1\ ~ ~ . I C,=D "\ / i ~ 'r-... CL = 0.1 Cl ~ 0.05 10 100 . 1k - J.LFj '\ '\ 10 '\ 10 J.LF rANT. 1.0 1 " ~ 0.1 V 10 k LOAD FREQUENCY - Hz 100 k 1M 0,0'10 o 500 .. 100 ,. FREQUENCY _ H2: 4-4 5k 10 k _21 0 25 so 75 100 CASE TEMPEAATURE _ OC '\ 125 110 ~A78H05 • ~A78H05A Typical Performance Curves (Cont.) Short Circuit Current 10 Quiescent Current Dropout Voltage Your" 5V > I '" 0-,. z>w I--.. '" i3'" !:: "" """'T 75·~'" """"'-.TJ -2S8 C J ~ = " " 0 iii o o .,0 20 30 3 i 2~--f---+---+---1---~ ~ ....... ~ ~---+--7'=-+---~-",,"-+---~ ffi M a 35 INPUT VOLTAGE _ V Line Regulation 'our- ~ 100 I > 'OUT-SA ~-10~---+-HL-+_---1-----r--~ w ~ i 1-20~---+-H--+---~-----r--~ " -~ ~-100 ~ -30~---+-H--+---~-----r--~ > g 5 ~-40~---+-H--+---~-----r--~ 5 I" 1 25 II ~ I ~ !rl OJ a: ~ 40 10 100 . ... 1 ~ Load Transient Response loUT ~ 14 IOUT l .our lSA 10 I 3A 1-20 I------+----+-----p.;~~;____l l'l ~-30 1~-100 \. r_+_t----1111 /'--t--t--+-+--t-+_~ ~-200 . ~-4-+~4-+-+-~-4 1M Output Voltage Deviation vs Junction Temperature V1N =10V i!l Ii ~ _50 ..... , I T= 2" = SA .....::: l'j ~ 1-1501--+----+-+--+-+--+--+-I 5 OUTPUT CURRENT _ A ~ I! -100 r_-+--+_-1-+--+--t-+~ ~-.or----t----+-----r----+----i -~OL--~--~~--~--~--~ lOY 100 1k 10k lOOk INPUT FREQUENCY _ Hz I 100 t-+-+--t-+--+-1-frCL = 0.1,.,.F ~-10~--_t~~~----r_--_+--__i ... '"'-"""" v ~ 2OOr-;--'--r-;--;-;~~~-'N~=~'0-Y-' > 1SO ~ .. 20 80 100 PULSE WIDTH nME _ 125 ·c Cl = O.1#tF I 100 ~ 20 7tii VIN .... " o ~ VIN = lOY 10 INPUT VOLTAGE _ V F""'" Ripple Rejection 120 i Load Regulation laJT=2A CL = 0.1,.,.F 20 - 1",.)= 3A r--- ~ JUNCTION TEMPERATURE - h'oUT=5A IIv 5 I lour'" 5A I-- INPUT VOLTAGE - Y Line Transient Response 2A I Ii I- o 20 40 10 PULSE WIDTH TIME - 4-5 eo It' 100 JUNCnOII T&IIPERATURE - 'C JLA78H05 • JLA78H05A Test Circuit Fixed Output Voltage 1 2 "A78H05 "A78H05A SOLID TANTALUM VOUT + + CIN 1 p.F 3 0.1 Cl T 1. Design Considerations These devices have thermal-overload protection from excessive power and internal short-circuit protection which limits the circuit's maximum current. Thus, the devices are protected from overload abnormalities. Although the internal power dissipation is limited, the junction temperature must be kept below the maximum specified temperature (150°C). It is recommended by the manufacturer that the maximum junction temperature be kept as low as possible for increased reliability. To calculate the maximum junction temperature or heat sink required, the following thermal resistance values should be used: Typ 8JC 1.8 Package TO-3 COMMON Caution: Permanent damage can result from forcing the output voltage higher than the input voltage. A protection diode from output to input should be used if this condition exists. Package Outline (S Package - Steel) .2.95 (7.49) .265 (6.73) r· 780 (19.81)1 .76001(A19.30) ~ SEATIN~ PlLA-NE-'---~~=* Max 8JC .450 (1 1.43) .400 (10.16) t 2.5 .057(1.45) .037 (0.94) .043 (1.09) .038 (0.97) TJ(max) - TA 8JC + 8CA Po (max) = 8CA =8cs + 8SA Solving for TJ: TJ = TA + Po (8JC + 8CA) .180 (4.57) A .150 (3.81) 2 PLACES .525 (13.34) A .480 (12. 19) PIN 1 Where: = Junction Temperature TJ TA = Ambient Temperature Po = Power Dissipation 8JC = Junction-to-case thermal resistance 8CA = Case-to-ambient thermal resistance 8CS = Case-to-heat sink thermal resistance 8SA Heat sink-to-ambient thermal resistance Notes All dimensions in inches bold and millimeters (parentheses) Pins are solder-dipped alloy 52 = The devices are designed to operate without external compensation components. However, the amount of external filtering of these voltage regulators depends upon the Circuit layout. If in a specific application the regulator is more than four inches from the filter capacitor, a 1 /LF solid tantalum capacitor should be used at the input. A 0.1 /LF capacitor should be used at the output to reduce transients created by fast switching loads, as seen in the basic test circuit. These filter capacitors must be located as close to the regulator as possible. 4-6 p;A78H12A 5-Amp Voltage Regulator FAIRCHILD A Schlumberger Company Hybrid Products Connection Diagram T0-3 Metal Package Description The IlA78H12A is a hybrid regulator with 12.0 V fixed output and 5.0 A output capability. It has the inherent characteristics of the monolithic 3-terminal regulators; i.e., full thermal overload, short-circuit and safe-area protection. All devices are packaged in hermetically sealed TO-3s providing 50 W power dissipation. If the safe operating area is exceeded, the device shuts down, rather than failing or damaging other system components (Note 1). This feature eliminates costly output circuitry and overly conservative heat sinks typical of high-current regulators built from discrete components. • 5.0 A OUTPUT CURRENT • INTERNAL CURRENT AND THERMAL OVERLOAD PROTECTION • INTERNAL SHORT CIRCUIT PROTECTION • LOW DROPOUT VOLTAGE (TYPICALLY 2.3 V@ 5.0 A) • 50 W POWER DISSIPATION • STEEL TO-3 PACKAGE (Top View) Order Information Type Package IlA78H12A Metal IlA78H12A Metal Note 1. This voltage regulator offers output transistor safe·area protection. However, to maintain full protection, the device must be operated within the maximum input-to·output voltage differential ratings, as listed on this data sheet under "Absolute Maximum Ratings." For applications violating these limits, device will not be fully protected. Part No. IlA78H12ASC IlA78H12ASM Code GN GN Block Diagram T T I START CIRCUIT CURRENT SOURCE VOLTAGE REGULATOR I ~t.l r- V" V IN THERMAL SHUTDOWN 1 1 SHORT CIRCUIT PROTECTION • R ...... "- K ...... Rsc II 2 VOUT 3 COMMON 4-7 JLA78H12A Absolute Maximum Ratings Input Voltage Input-to-Output Voltage Differential, Output ShortCircuited Internal Power Dissipation Operating Junction Temperature Military Temperature Range ILA78H12ASM ILA7812A Electrical Characteristics 40V Commercial Temperature Range ILA78H12ASC Storage Temperature Range Pin Temperature (Soldering, 60 s) 35 V 50 W @ 25°C Case O°C to +150°C -55°C to +150°C 300°C 150°C -55°C to +150°C TJ = 25°C, VIN = 19 V, lOUT = 2.0 A unless otherwise specified Limits Symbol Characteristic Condition Min Typ Max Unit VOUT Output Voltage lOUT 11.5 12 12.5 V AVOUT Line Regulation (Note 2) = 2.0 A VIN = 16 to 25 V 20 120 mV AVOUT Load Regulation (Note 2) 10 mA ::; lOUT::; 5.0 A 20 120 mV 10 Quiescent Current 10 mA RR Ripple Rejection Vn Output Noise = 0, VIN = 17 V lOUT = 1.0 A, f = 120 Hz, 5.0 Vpk-pk 10 Hz ::; f ::; 100 kHz, VIN = 17 V lOUT = 5.0 A lOUT = 3.0 A 3.7 lOUT Voo Dropout Voltage (Note 3) loS Short-Circuit Current Limit Notes 2. Load and line regulation are specified at constant junction temperature. Pulse testing is required with a pulse width :$ 1 ms and a duty cycle :$ 5%. Full Kelvin connection methods must be used to measure these parameters. 60 dB 75 2.3 2.5 VRMS V 2.0 2.3 V 7.0 12.0 Apk 3. Dropout Voltage is the input·to-output voltage differential that causes the output voltage to decrease by 5% of its initial value. Typical Performance Curves Output Impedance Output Noise Voltage 100 0, Maximum Power Dissipation 1. louT = 1A VIN ... 11V 50 YIN"" c" r0- 0.5 Ct.=o "\ = 0 C, =0.1"i/ 10,., 1,\ 1,\ 10 ,ANT. 1.0 10 1\ r1" / 1 '\ 1\ I- ) t-..... i 50 r---. 1 C, 80 'IV ,. 1k 10k LOAD FREQUENCY _ Hz 1DOic 111 0.01 o 10 500 1 Ie 50 100 FREQUENCY - Hz 4-8 Sic 10k -25 0 25 50 75 100 CASE TEMPERATURE - ·C 128 150 ,uA78H12A Typical Performance Curves (Cont.) Quiescent Current Short Circuit Current Dropout Voltage • 12 c" ~ TJ = -" .. 3 r---.:: t'•1. ~J=12'.C TJ = 25°C "" TJ = 25°C TJ~ r:::--- 2 / ,.- .& 2. 4. 3. INPUT VOLTAGE - 1 ) I 2. 20 " ~ IOUT!2A IOUT=5A 100 50 ...> ~- IOUT=SA I Iv "z. '" !; U ,. INPUT . ,. VOLT~GE Load Regulation ~ I -10~---t~~~--~-----r----; ~ l'l 5 >-20~---t----+---~~~~~--; o .. ~ 20 40 60 200 IOUT . 2. 80 100 125 150 CL "" ,IY 0.1 p.F l 3A IQurlSA I 1 10 100 1k '"" ~ ~V 10 k 100 k 1M INPUT FREQUENCY - Hz Output Voltage Deviation vs Junction Temperature YIN'" 19Y '00 ~+-t--t-+--+~---r.-CL " = 0.1 IJ.F ~-200~+-+~t-+--+~--+--r~--1 ~ 1----+----+----'-1----,...__""""1 ~ ~ f---f---f---f---'-r-----l -100 ~-+--+--f--t--+-~---r--; 1-'50 f-----+--+----+--+--+--+---+---l o OUTPUT CURRENT - A 100 ~~_ 100 ~+-+----'IIIr,--+--+-+-+--f-+---l !; !; ~ -4. ~ ~ Load Transient Response Ie §! eo PULSE WIDTH TIME - P.s I ~ ~ -30 Z ;;J a: o ~ 75 ~ I 10 - V 50 I ill 2. 2. 3Ar-- YIN = ;!; 20 2. 100 II w 0 • _25 =0 0 > - P- 1 YIN 19 Y CL = O.lIJ.F II . -50 II lou; lOUT = 2A Ripple Rejection l'l 0 i ~ 120 r'\ i!i Ii:;-20 ~-30 lOUT-SA JUNCTION TEMPERATURE _ ·C Line Transient Response E I -10 -' INPUT VOLTAGE - V V Line Regulation > I " ,. 00 3 > 20 40 60 80 100 PULSE WIDTH nMe - /1-. 4-9 JUNCTION TEMPERATURE _ ·C #L A78H12A Basic Test Circuit 1 2 )lA78Hi2A SOLID TANTALUM VOUT + + C1N l/LF i' 3 CL 0.1 /L F COMMON .1 Design Considerations Caution: Permanent damage can result from forcing the output voltage higher than the input voltage. A protection diode from output to input should be used if this condition exists. This device has thermal-overload protection from excessive power and internal short-circuit protection which limits the circuit's maximum current. Thus, the device is protected from overload abnormalities. Although the internal power dissipation is limited, the junction temperature must be .kept below the maximum specified temperature (150°C).-lt is recommended by the manufacturer that the maximum junction temperature be kept as low as possible for increased reliability. To calculate the maximum junction temperature or heat sink required, the following thermal resistance values should be used: Package Outline (S Package - Steel) .295 (7.49) .265 (6.73) ~ SEATlN~ r· 780 (19.81)-1 .76°ot(A19.30) .057(1.45) .037 (0.94) PLJ...A-NE----'"r-t...,.,=* Package TO-3 PO(max) = Typ Max (JJC (JJC 1.8 2.5 .450 (11.43) .400 (10.16) * TJ(max) - TA (JJC + (JCA .161 (4.09) otA .151 (3.84) 2 HOLES = (JCS + (JSA (JCA Solving for TJ: TJ = TA + Po «(JJC .043 (1.09) .038 (0.97) + (JJA) Where: TJ Junction Temperature TA Ambient Temperature .po Power Dissipation (JJC Junction-to-case thermal resistance (JCA Case-to-ambient thermal resistance (JCS Case-to-heat sink thermal resistance (JSA = Heat sink-to-ambient thermal resistance = = = Notes All dimensions in inches bold and millimeters (parentheses) Pins are solder-dipped alloy 52 = = = The devices are designed to operate without external compensation components. However, the amount of external filtering of these voltage regulators depends upon the circuit layout. If in a specific application the regulator is more than four inches from the filter capacitor, a 1 IlF solid tantalum capacitor should be used at the input. A 0.1 IlF capacitor should be used at the output to reduce transients created by fast switc::hing loads, as seen in the basic .test circuit. These filter capacitors must be located as close to the regulator as possible. 4-10 JlA78HGA Positive Adjustable 5-Amp Voltage Regulator FAIRCH.ILD A Schlumberger Company Hybrid Products Description The ~A78HGA is an adjustable 4-terminal positive voltage regulator capable of supplying in excess of 5.0 A over a 5.0 V to 24 V output range. Only two external resistors are required to set the output voltage. Connection Diagram TO-3 Metal Package The ~A78HGA is packaged in a hermetically sealed TO-3, providing 50 W power dissipation. The regulator consists of a monolithic chip driving a discrete seriespass element. A beryllium-oxide substrate is used in conjunction with an isothermal layout to optimize the thermal characteristics of each device and still maintain electrical isolation between the various chips. This unique circuit design limits the maximum junction temperature of the power output transistor to provide full automatic thermal overload protection. If the safe operating area is ever exceeded (Note 1), the device simply shuts down rather than failing or damaging other system components. This feature eliminates the need to design costly regulators built from discrete components. (TOp View) • 5_0 A OUTPUT CURRENT • INTERNAL CURRENT AND THERMAL LIMITING • INTERNAL SHORT CIRCUIT CURRENT LIMIT • LOW DROPOUT VOLTAGE (TYPICALLY 2.3 V@ S.OA) • 50 W POWER DISSIPATION • ELECTRICALLY NEUTRAL CASE • STEEL TO-3 PACKAGE • ALL PIN-FOR-PIN COMPATIBLE WITH ~A78HG Order Information Type Package ~A78HGA Metal ~A78HGA Metal Code Part No. JA JA ~A78HGASC ~A78HGASM Block Diagram-Positive Adjustable Voltage Regulator I I START CIRCUIT I CURRENT SOURCE I VOLTAGE REGULATOR I ~~ ,..... ;:/' VIN THERMAL SHUTDOWN R 1'- ..... ..... 1 1 SHORT CIRCUIT PROTECTION L ,...,., ~ Rsc 2 VOUT 3 CONTROL 4 COMMON Notes on following pages. 4-11 ILA78HGA Absolute Maximum Ratings Input Voltage Internal Power Dissipation Maximum Input-to-Output Voltage Differential Output Short Circuit Operating Junction Temperature Military Temperature Range Il A7 8HGASM Electrical Characteristics 40 V 50 W @ 25°C Case Commercial Temperature Range Il A78HGASC Storage Temperature Range Pin Temperature (Soldering, 60 s) 35 V O°C to +150°C -55°C to +150°C 300°C 150°C -55°C to +t50°C TJ = 25°C, VIN = 10 V, lOUT = 2.0 A unless otherwise specified Limits Typ Symbol Characteristic Condition (Note 3) Min VOUT Output Voltage (Note 4) lOUT = 2.0 A, VIN = VOUT + 3.5 V 5.0 ~VOUT Line Regulation (Note 2) VIN = 7.5 to 25 V 0.2% ~VOUT Load Regulation (Note 2) 10 mA :5 lOUT :5 5.0 A 10 Quiescent Current lOUT = 0 RR Ripple Rejection lOUT = 1.0 A, f = 210 Hz, 5.0 Vpk-pk 60 Vn Output Noise 10 Hz :5 f :5 100 kHz, VIN= VOUT + 5.0 V 50 VOO Dropout Voltage (Note 5) lOUT = 5.0 A 2.3 2.5 V lOUT = 3.0 A 2.0 2.3 V lOS Short-Circuit Current Limit Vc Control Pin Voltage Unit 24 V 1% V 0.2% 1% V 3.4 10 mA dB VIN = 15 V 4.85 Notes 1. This voltage regulator offers output transistor safe-area protection. However, to maintain full protection, the device must be operated within the maximum input-to-outpufvoltage differential rating listed on the data sheet under •. Absolute Maximum Ratings." For applications violating these limits, device will not be fully protected. 2. Load and line regulation are specified at constant junction temperature. Pulse testing is required with a pulse width !S 1 ms and a duty cycle !S 5%. Full Kelvin connection methods must be used to measure these parameters. Max Il VRMS 7.0 12.0 Apk 5.0 5.25 V 3. The performance characteristics of the adjustable series (I'A78HGA) is specified for VOUT ,. 5.0 V, unless otherwise noted. Rl + R2 4. VOUT is defined as VOUT = (VCONT) where Rl and R2 are defined in the Basic Test Circuit diagram. 5. Dropout Voltage is the input-output voltage differential that causes the outp.ut voltage to decrease by 5% of its initial value. R2 Typical Performance Curves Output Noise Voltage Output Impedance Maximum Power Dissipation so 1.0 100 lOUT - 1 A YOUT = 5 V so 10V V'N 0.5 YIN'" tOY "'r-. YOUT = 5V CL 0 0 z r--., C L = 0 ""\ 1 ) """" r---. ./ Tl' CL -o.1,...F/ 2.0 CL 10 I~ II: ~ 0 2 1\ 1'\ 0 1 i\ 10 /L f rANT. 1.0 1 ~ 40 100 1k 10 k LOAD FREQUENC'( _ Hz 100 k 1M 0.01 0 10 so 100 soo ,. FREQUENCY - Hz 4-12 5k 10 k -25 0 25 50 75 100 CASE TEMPERATURE _ °C [i\ 125 150 JLA78HGA Typical Performance Curves (Cant.) • 10 TJ= c E """,TJ = 25°C ........ TJ =15"C,," I~ ...... .- 2 o o • ~ i I .f---- 30 20 "TJ = " .. ~ II e TJ=25-C 1 I I louT = SA I T "T IOUT=3A 1 20 15 10 25 i 2. INPUT VOLTAGE - V INPUT VOLTAGE _ V f-- IOUT=2A ~ !! 35 75 50 100 121 150 JUNCTION TEMPERATURE - OC Ripple Rejection Line Transient Response Line Regulation VOUT =5V > I 2 0/ 810 -lex ~ ...~ I J'... t'--, Dropout Voltage Quiescent Current Short Circuit Current 120 > IIV 'OUT=5A ~-10~--+-H~+---1--~---1 lou! 5AI= ,louT=3A ~ L20I--+-H--I--+---iI---I CL = 0.1 p.F 100 ~J V ~ -30~~-+-H-+---1--~---1 I Z 0 ~ 80 T t 1~3A 5 50 II: g ~ ~ 5 $-~r-~-&~---+---+--~ 'curlsA ~ o 20 eo 80 PULSE WIDTH TIME - INPUT VOLTAGE - Y Load Regulation 40 100 I 1 10 ~ ~ 200 VIN = 10V I 100 1--+-t--t--+--+--j--j.frVOUT '" 5 V I~L 100 1k ''"'""" 10 k 1M 50 VIN = 10Y rOUT = 0.1 p.F 5 $-200 c 7 100 k Output Voltage Deviation vs Junction Temperature i1.>-100 1-+-+-4I/-/f-+--+--+-1~+--I o ~ INPUT FREQUENCY - Hz Load Transient Response III 20 YIN = lOY YOUT = 5V CL = O.1IJ.F _ I Your = SV ~ f-- f!1ouT=IA i!l VIN = lOY I'V 7UT 12A p In, T == SA 1-+-+-+-+-1--~+-4-~~ F::::::: I ! I.> -500L__ ~_~_-~---L---~ OUTPUT CURRENT _ A I o 20 40 80 80 100 PULSE WIDTH TIME - lAS 4-13 -200 -SO -25 25 50 75 100 JUNCTION TEMPERATURE _ ·C 125 150 J-LA78HGA Test Circuit Adjustable Output Voltage 2 VOUT 1 ~A78HGA C IN = 21-'F :::::: + SOLID TANTA LUM Rl CONTROL + 3 :::::: CL'" 1.0 I-' F ~ 4 R2 COMMO N Caution: Permanent damage can result from forcing the output voltage higher than the input voltage. A protection diode from output to input should be used if this condition exists. Design Considerations This device has thermal-overload protection from excessive power and internal short-circuit protection which limits the circuit's maximum current. Thus, the device is protected from overload abnormalities. Although the internal power dissipation is limited, the junction temperature must be kept below the maximum specified temperature (150°C). It is recommended by the manufacturer that the maximum junction temperature be kept as low as possible for increased reliability. To calculate the maximum junction temperature or heat sink required, the following thermal resistance values should be used: Package TO-3 Voltage Output The device has an adjustable output voltage from 5.0 V to 24 V which can be programmed by the external resistor network (potentiometer or two fixed resistors) using the relationship R1 + R2) VOUT = VCONTROL ( R2 Example: If R 1 = 0 f! and R2 = 5 kf!, then VOUT = 5 V nominal. Or, if R1 = 10 kf! and R2 = 5 kf!, then VOUT = 15 V. Max liJC 2.5 Typ liJC 1.8 Package Outline (S Package - Steel) TJ(max) - TA PO(MAX) liCA = li JC + liCA = lics + liSA Solving for TJ: TJ TA + Po (liJC = ~ ~I + liCA) .293 (7 44) .273 (6 93) Where: Junction Temperature TJ Ambient Temperature TA Po = Power Dissipation liJC = Junction-to-case thermal resistance liCA Case-to-ambient thermal resistance liSA = Heat sink-to-ambient thermal resistance liCS Case-to-heat sink thermal resistance = = .421 (10.69>! MIN _ r Ii· ~:~~~g::~~.770(1956) MAX f-- I 1 n n UU III-, I + : I i 1- t SEATING PLANE .067 (170) MAX I __ .041 (1.04) DIA .037 (940) . ___ 1.197 (30.40) ___ 1.177 (29.90) = = PIN 1 I----~~ :~~~ ~ ~~~:~ PIN 2 This device is designed to operate without external compensation components. However, the amount of external filtering of this voltage regulator depends upon the circuit layout. If in a specific application the regulator is more than four inches from the filter capacitor, a 1 p.F solid tantalum capacitor should be used at the input. A 0.1 p.F capacitor should be used at the output to reduce transients created by fast switching loads, as seen in the basic test circuit. These filter capacitors must be located as close to the regulator as possible. 2 HOLES .161 (409) .151 (3.84) .177 (45) R 2 PLACES .470 (11.94) DIA PIN CIRCLE Notes PIN 3 .525 (1334) MAX All dimensions in inches bold and millimeters (parentheses) 4-14 J,LA78P05 5-Volt 10-Amp Voltage Regulator FAIRCHILO A Schlumberger Company Hybrid Products Connection Diagram TO-3 Metal Package Description The ILA78P05 3-terminal positive 5 V regulator, consisting of a monolithic control chip driving a seriespass transistor, is capable of delivering 10 A. This hybrid device is virtually blow-out proof and contains all the protection features inherent in monolithic regulators such as internal short-circuit current limiting, thermal overload and safe-area protection. If the safe-operating area is exceeded, the device shuts down rather than failing or damaging other system components (Note 1). This feature eliminates costly output circuitry and overly conservative heat sinks typical of high-current regulators built with discrete components. The ILA78P05 is packaged in a hermetically sealed TO-3 providing 70 W power dissipation. • • 10 A OUTPUT CURRENT • INTERNAL THERMAL OVERLOAD PROTECTION • INTERNAL SHORT CIRCUIT CURRENT LIMIT • LOW DROPOUT VOLTAGE (TYPICALLY 2.3 V @ 10 A) • 70 W POWER DISSIPATION • PIN-FOR-PIN COMPATIBLE WITH THE ILA78H05, ILA78H05A AND SH323 • STEEL TO-3 PACKAGE (Top View) Order Information Type Package ILA78P05 Metal ILA78P05 Metal Note 1. This voltage regulator offers output transistor safe-area protection. However, to maintain full protection, the device must be operated within the maximum input·to-output voltage differential ratings as listed on this data sheet under "Absolute Maximum Ratings." For applications violating these limits, device will not be fully protected. Part No. ILA78P05SC ILA78P05SM Code 6N 6N Block Diagram I l START CIRCUIT I CURRENT SOURCE VOLTAGE REGULATOR I ~tL ~ ;'/' VIN THERMAL SHUTDOWN 1 R ...... ...... ,.... V I SHORT, CIRCUIT PROTECTION ~ Rsc I} 2 VOUT 3 COMMON 4-15 ~A78P05 Absolute Maximum Ratings Input Voltage Input-to-Output Voltage Differential, Output ShortCircuited Internal Power Dissipation Operating Junction Temperature Military Temperature Range JLA78P05SM Commercial Temperature Range JLA78P05SC Storage Temperature Range Pin Temperature (Soldering, 60 s) 40 V 35 V 70 W @ 25°C Case 150°C JLA78P05 Electrical Characteristics TJ = 25°C, VIN = 10 V, lOUT = O°C to +150°C -55°C to +150°C 2.0 A unless otherwise specified Limits Symbol Characteristic Condition Min Typ Max Unit VOUT Output Voltage lOUT 4.85 5.0 5.25 V ~VOUT Line Regulation (Note 2) = 2.0 A VIN = 8 to 25 V 10 50 mV ~VOUT Load Regulation (Note 2) 10 mA ::S lOUT ::S 5 A 25 40 mV ~VOUT Load Regulation (Note 2) 10 mA ::S lOUT ::S 10 A 50 75 mV IQ Quiescent Current lOUT 3.4 10 mA RR Ripple Rejection =0 lOUT = 1.0 A, f = 120 Hz, 5.0 Vok-pk Vn Output Noise VDD Dropout Voltage (Note 3) loS Short-Circuit Current Limit 60 dB 10 Hz ::S f ::S 100 kHz 40 lOUT = 5.0 A 2.0 2.3 V = 10 A 2.5 3.0 V lOUT JLVRMS 14 Notes 2. Load and line regulation are specified at constant junction temperature. Pulse testing is required with a pulse width :oS 1 ms and a duty cycle :oS 5%. Full Kelvin connection methods must be used to measure these parameters. Apk 3. Dropout Voltage is the input-output voltage differential that causes the output voltage to decrease by 5% of its initial value. Typical Performance Curves Output Noise Voltage 1.0 0.5 ~ v" II 1-- - ~ e'l --- - r-l"- r-l f'.- I 0.1 f-0.01 - tt-- m l- t- -- : I 1- - + 'II I 10 50 100 500 1k FREQUENCY - Hz 70 5 k 10 k r\ IOUT=1A V1N =10V 60 50 ~ I f\. 50 z 20 CL '! He- f--j Maximum Power DisSipation 100 0 ~$ I 0.05 Output Impedance lOV '" i 0 ....... 10 V !II c )t--... 5.0 2.0 CL 110 ~ it /A.Fj 100 1k 10 k LOAD FREQUENCY - Hz 4-16 r\ 20 \ 10 o 10 :\ 30 \ /l-F rANT. 1.0 1 \ 0: / C L - 0.1 40 100 k 1M -25 0 25 50 75 100 CASE TEMPERATURE _ G 125 C 150 JLA78P05 Typical Performance Curves (Cent.) Short Circuit Current ,. c .. ,. " ,. Quiescent Current I TJ= I- Z ~r--., ~ U -.. TJ - 2S"C "0t: "!Iiu TJ - 75°C 4 ~ 3 , I- ~ C E r--- ~ ~ 10 I ... u 0 ill •o .,0 20 INPUT VOLTAGE - V "~ IOU)'- SA- l - - I ·20 z o o i:i~ -40 . !; - lOUT·· 10 A o g 1. 10 •0 •• I' -60 ~ o" 10 20 15 It~,I,- -~ ,i Load Regulation _1001--+-+--"F'O,U"-'_"i'S-=A+-+_tl->+---t--t -2001---+-+-I---+-+-f---+-+-t-l o I -50 z --- o ~ ~ .. T, 20 40 60 ~ _25°C ~S·C~ ....... ~ T, 1~ -100 o ~ "~ 100 0.4 '--""""'-'--"""""''''''-'-V-'Nr-_-,-'0-V' O.21--+-+-I--+-+--t-f,.-CL =- 0.1 "F I\, ~ .O.• I---+-+--Iff--+-+--I-+-+-+-I i -O.41---+-+---'f---+-+-t-+-+-+-l !; o 125 150 °C VIN :: ..", Z 0 .~ 100 ~OV Ct. = 0.1 ~F IOUT~'A .. ~ IoUT l 1'\ 3A 60 . loUT ls "7 A '" I 20 80 100 o 1 10 100 1k 10 k ~ 1M 100 k INPUT FREQUENCY - Hz Output Voltage Deviation vs Junction Temperature 50 YIN=10Y ~ i. - IO~T - {o A I r-- -.:r- ~ IOUT=2A -so ......... ~ "~ -150 ~ -100 g I- !; !; o S ~ -200 I!: -150 VIN -250 75 I PULSE WIDTH TIME-pi Load Transient Response T, so 120 ~ INPUT VOLT AGE - V ~ .5 ;;S •• ,A Ripple Rejection ~ o '0 , JUNCTION TEMPERATURE - ~ -80 "o -100 o -'S 200,-...,.......,.-.--........,.-.--,......,.......,-, ~ 100 1--+-+-SJFj1'_OU_'+=_'_0I-A-+ ~~N _.~ O~~ :F lOUT - 2 A 'OU~!f I ~ > E A- 1 Line Transient Response 0 ~ -F=:: TJ = 25°C INPUT VOLTAGE - V Line Regulation ~ ) ,J, -.J ""- "'TJ = , ••·C ( 00 3. 30 1 J. c, ~ ~ , . Dropout Voltage o I 10 V 10 OUTPUT CURRENT - A o 20 40 80 80 100 PULSE WIDTH TIME - ..... 4-17 -200 -50 -25 25 50 75 100 125 JUNCTION TEMPERATURE _ °C 150 jlA78P05 Basic Test Circuit 2 1 "A78P05 SOLID TANTALUM + + CIN CL 3 0.1 /L F l/L F COMMON ~ Design Considerations This device has thermal-overload protection from excessive power and internal short-circuit protection which limits the circuit's maximum current. Thus, the devices are protected from overload abnormalities. Although the internal power dissipation is limited, the junction temperature must be kept below the maximum specified temperature (150°C). It is recommended by the manufacturer that the maximum junction temperature be kept as low as possible for increased reliability. To calculate the maximum junction temperature or heat sink required, the following thermal resistance values should be used: Caution: Permanent damage can result from forcing the output voltage higher than the input voltage. A protection diode from output to input should be used if this condition exists. Package Outline (S Package - Steel) .295 (7.49) .265 (6.73) ~ SEATIN~ r· 780 (19.81)1 .760DI(A19.30) .057(1.45) .037 (0.94) PL-'-A-N-E----'-r-~ ~ I 7"C Z .. '\ 1\ I", 4. II 2.~ -6 -4 U L2 •o ,. -5 -10 -15 Dropout Voltage Quiescent Current -20 _25 -30 -35 • -25 0 25 50 75 INPUT VOLTAGE - V lOUT '" '" 100 CASE TEMPERATURE _ °C 4-20 125 150 = -2A .~~--~~--~~--~~ -25 0 25 50 75 100 125 JUNCnON TEMPERATURE - QC 150 f.LA79HG Typical Performance Curves (Cont.) Line Transient Response ~ Load Regulation I-+-+---jf-+-+-+Vour= -s.oy lOUT = -3.0 A 20 i w 10~+-+~f-+-+-1--+-~-+~ ~ ol-+-+....,f--t--t-j--i?+-+--I ~-101-+-+---j~+-+~--f-~-+-4 ~ ~-20~+-+-1f-+-+-1__+-~-+~ > ~-101-+-+f-f--t--t-j--~+-+--I !1 !:; g-~I-+-+f-f--t--t-j--+-+-+--I i 1- 10 25'C VIN=-10Y \ "-75"C \. 125-C ~ Ie > -20 i!! w ~ "g -30 5 1!:-40 5 -50 PULSE WIDTH TIME - Vo!.T ='-5.~V 1\ '- 'E Load Transient Response o _1 ~8 -2 -3 -4 -5 • OUTPUT CURRENT - A Output Voltage Deviation vs Junction Temperature Control Current vs Temperature Differential Control Voltage vs Input Voltage 0.7 0 .• VIN = -40V ~ ~ Ie -2A -5A ~w -50 ~-+---I----1f-+--+--t--+~ ~ "g !Ew '"'" '" 0.2 ~ ~ ]'\ YIN L-L__....L..__.L.-L__..L__' - - ' - _ -25 25 50 75 100 125 JUNCTION TEMPERATURE _ ac 150 Maximum Power Dissipation -5 ~ ~ -4 1-_ 25";"- / u ffi -2 ,,~ -1 o o -5 ~ ~ ~ ~ 75"C -10 -15 -20 -25 -30 -35 -40 INPUT VOLTAGE - V o o I' C'-... TJ=25"C,", '~UTi 'fmf 25 50 75 i' ~ <1- 8.0 "r-- = 10 v = -5.0 Y YOUT I' !iE-I.a I"'!, 0.1 _200 t-... > -4.0 E I"'!, 0._ 6 5 -150 f---+---I--I---I--I---r---jf--- r--- r--- 0.5 " U -2.0 OA 5 PULSE WIDTH 11ME - Id 100 -10 "r-. 125 JUNCTION TEMPERATURE _ °C 150 lour: -12 -5.0 _2 A YOUT = -S.OY -10 -15 -20 -25 INPUT VOLTAGE - V -30 j.LA79HG Basic Test Circuit, Adjustable Output Voltage 3 VOUT 4 2~F 1 C IN = SOLID TANTA LUM ~A79HG + R1 CONTROL + 2 CL'" 1.0 ~ F ~ 1 R2 VOUT COMM ON (R1 + R2) = VCONT'--R-2- Caution: Permanent damage can result from forCing the output voltage higher than the input voltage. A protection diode from output to input should be used if this condition exists. Design Considerations This device has thermal overload protection from excessive power and internal short circuit protection which limits the circuit's maximum current. Thus, the device is protected from overload abnormalities. Although the internal power dissipation is limited, the junction temperature must be kept below the maximum specified temperature (150°C). It is recommended by the manufacturer that the maximum junction temperature be kept as low as possible for increased reliability. To calculate the maximum junction temperature or heat sink required, the following thermal resistance values should be used. Package Typ Max TO-3 IJJC 1.8 IJJC 2.5 Voltage Output The device has an adjustable output voltage from -2.11 to -24 V which can be programmed by the external resistor network (potentiometer or two fixed resistors) using the relationship: VOUT = VCONTROL Package Outline (S Package - Steel) + IJSA Solving for TJ: TJ = TA + Po (lJJC + R2) R2 Example: If R 1 = 0 f! and R2 = 5 kf!, then VOUT = -2.11 V nominal. Or, if R1 = 12.8 kf! and R2 = 2.1 kf! then VOUT = -15 V. TJ(MAX) - TA PO(MAX) = IJJC + IJCA IJCA = IJcs ( R1 + IJCA) .293(744) .273 (6 93) _I J± r I _ _ Ii 1 Where: TJ = Junction Temperature TA = Ambient Temperature Po = Power Dissipation IJJC = Junction-to-case thermal resistance IJCA = Case-to-ambient thermal resistance IJcs = Case-to-heat sink thermal resistance IJSA = Heat sink-to-ambient thermal resistance i .421 (1069) MIN ~ _ PIN 1 ~;;~ g~ :~;~I .770 (1956) MAX I1 I n n U U _IIt - I + SEATING PLANE : : : 1-·- i .067 (170) MAX .041 (104) DIA .037 ( 940) 1.197 ( 3 0 4 0 ) _ 1.177 (2990) I-------<~ .675 (17 14) .655 (1664) PIN 2 2 HOLES .161 (409) .151 (384) The device is designed to operate without external compensation components. However, the amount of external filtering of these voltage regulators depends upon the circuit layout. If in a specific application the regulator is more than four inches from the filter capacitor, a 2 ,uF solid tantalum capacitor should be used at the input. A 1 ,uF capacitor should be used at the output to reduce transients created by fast switching loads, as seen in the basic test circuit. These filter capacitors must be located as close to the regulator as possible. .177 (4 5) R 2 PLACES .470 (1194) DIA PIN PIN 3 CIRCLE .525 (1334) MAX Notes All dimensions in inches bold and millimeters (parentheses) Pins are solder-dipped alloy 52 4-22 SH323 • SH223 • SH 123 3 A, 5 V Voltage Regulator· FAIRCHILD A Schlumberger Company Hybrid Products Connection Diagram 2-Pin Metal Package Description The SH323 is a hybrid regulator with 5.0 V fixed output and 3.0 A output capability. It has the inherent characteristics of the monolithic 3-terminal regulators, Le., full thermal overload, short circuit and safe area protection. All devices are packaged in hermetically sealed TO-3s providing 50 W power dissipation. If the safe operating area is exceeded, the device shuts down ·rather than failing or damaging other system components (Note 1). This feature eliminates costly output circuitry and overly conservative heat sinks typical of high-current regulators built from discrete components. • 3.0 A OUTPUT CURRENT • INTERNAL CURRENT AND THERMAL OVERLOAD PROTECTION • INTERNAL SHORT CIRCUIT PROTECTION • LOW DROPOUT VOLTAGE (TYPICALLY 2.0 V @3.0A) • 50 W POWER DISSIPATION • STEEL TO-3 PACKAGE • ALL PIN-FOR-PIN COMPATIBLE WITH THE LM323, SG323 (Top View) Order Information Type Package SH323 Metal SH223 Metal SH123 Metal Part No. SH323SC SH223SV SH123SM Code GN GN GN Block Diagram T T I START CIRCUIT CURRENT SOURCE VOLTAGE REGULATOR I e-~ r- ~ THERMAL SHUTDOWN VIN .... 1 1 SHORT CIRCUIT PROTECTION ,, R ~ K ...... Rsc IJ 2 VOUT 3 COMMON 4-23 SH323 • SH223 • SH 123 Absolute Maximum Ratings Input Voltage Input-to-Output Voltage Differential Output Short Circuited Internal Power Dissipation Operating Junction Temperature Industrial Temperature Range SH223SV Electrical Characteristics TJ Military Temperature Range SH123SM Commercial Temperature Range SH323SC Storage Temperature Range Pin Temperature (Soldering, 60 s) 40 V 35V 50 W @ 25°C Case 150°C -55°C to +150°C O°C to +150°C -55°C to +150°C 300°C -25°C to +150°C '. = 25°C, VIN = 10 V, lOUT = 2.0 A unless otherwise specified. Limits Symbol Characteristic Min Typ Max Unit Condition VOUT Output Voltage 4.85 5.0 5.25 V lOUT .:lVOUT Line Regulation (Note 2) 10 25 mV VIN .:lVOUT IQ RR Load Regulation (Note 2) 10 3.0 50 mV 10 mA ::5 lOUT ::5 3.0 A 10 mA lOUT Ripple Rejection dB lOUT Vn Output Noise 40 VOO Dropout Voltage (Note 3) 2.0 2.3 V los Short Circuit Current Limit 7.0 12.0 Apk Quiescent Current 60 ~VRMS Notes 1. This voltage regulator offers output transistor safe area protection. However, to maintain full protection, the device must be operated within the maximum input·to·output voltage differential ratings, as listed on this data sheet under" Absolute Maximum Ratings." For applications violating these limits, device will not be fully protected. 2. Load and line regulation are specified at constant junction = 2.0 A = 7.5 to 25V =0 = 1.0 A, f = 120 Hz, 5.0 Vpk-pk 10 Hz ::5 f ::5 100 kHz, VIN = 10 V lOUT = 3 A VIN = 10 V temperature. Pulse testing is required with a pulse width :os 1 ms and a duty cycle :os 5%. Full Kelvin connection methods must be used to measure these parameters. 3. Dropout Voltage is the input·output voltage differential that causes the output voltage to decrease by 5% of its initial value. Typical Performance Curves . ,. Short Circuit Current . . I '-.. "" .,. 1\ ~TJ=25"C TJ =75-e,,", • > 50 ......... a Dropout Voltage Maximum Power Dissipation ......... i"-. ao INPUT VOLTAGE _ V '\ ....... K 30 '\ ,. 35 • -25 0 2S 5Q 75 II 1,\ 1,\ 100 CASE TEMPERATURE _ ·C 4-24 125 150 'oJT=31 lour=2A :::;r ~ I• -as 25 50 75 100 JUNCTION TEMPERATURE - ·C 125 150 SH323 • SH223 • SH 123 Typical Performance Curves (Cont.) Line Regulation Line Transient Response Ripple Rejection 120 ~I VIN"" 10V CL ", 0.1 J-LF lOUT'" 2 A -10 f---+-+-+-"--1'---+--j :; -20 f---+-+-+-"--1--+--j 100 ~ ~ !li !;i i!lw I Z 0 80 ;:: ~ g l;l OJ f---+-+--f--"--1--+--j -30 a: ~ 5~ -40 r----+-ir-+---+---f----j 10 15 20 ~ o 25 Load Regulation ........... ~ -10 t'- .1 ..1 ~ / T J "" ', 2So C -~ 10 ",5 ~ 1/ o " g~ -100 ~-+-i--~-+-i--~-t-, :J: 5 o~ -150 20 40 60 1.0 lOUT = 1 A V1N =10V £! "> r- '> < I"-- I V ~ )r--. g n- . / CL = O.l.F / 0.1 1---- I 1 10 100 1k 10 k LOAD FREQUENCY - Hz I 100 k 1M 0.01 Quiescent Current VIN 10 V C 0 += lS r- 10 I I C l '" 10 ~F TANT. 1.0 JUNCTION TEMPERATURE - ·C I ~ 0.05 i f---t---t--t---t---t--t---t--j J- 1 I"-- 0.5 CL = 0 " r- _ 200 I--'-_-'-_'--'-_-'-_'---'-_...J _ 50 - 25 25 50 75 100 125 150 80 100 Output Noise Voltage 100 r:; w o 20 IOUT=2A w PULSE WIDTH TIME - /.LS Output Impedance ~ t( t----tr-:;;;-I--t--t-"*=::::::r---r-1 OUTPUT CURRENT - A / 1M -50 § r-- lOOk o ~ - 10k O "5 r--- 1k VOUT vs Junction Temperature I--+--+-+--+-+-f--t CL "" 0 1 I-tF I' -50 1--+--+-1/1/'---1-+-+--+-+--+--1 ~ -100 1--+--+-t-+--t-f--f-+--t---1 :> ~_40 2.0 100 INPUT FREQUENCY - Hz I 10 ~ 50 w TJ '" 2S·C ---' 5.0 " . 20 80 100 il 5" 5 -50 60 ~ VIN == 10V YIN"" 10V ~-25·C~ ~~ -30 g 40 Load Transient Response ~ !:i;;: -20 i!l 20 PULSE WIDTH TIME INPUT VOLTAGE _ V I 10UT = 3 A 60 50 100 1 11 500 1k FREQUENCY - Hz 4-25 5 k 10 k INPUT VOLTAGE - V • SH323 • SH223 • SH 123 Test Circuit Fixed Output Voltage 1 2 SH323 SOLID TANTALUM VOUT + + r CL 0.1 3 l/'F CIN COMMON ~ Design Considerations This device has thermal overload protection from excessive power and internal short circuit protection which limits the circuit's maximum current. Thus, the device is protected from overload abnormalities. Although the internal power dissipation is limited, the junction temperature must be kept below the maximum specified temperature (150°C). It is recommended by the manufacturer that the maximum junction temperature be kept as low as possible for increased reliability. To calculate the maximum junction temperature or heat sink required, the following thermal resistance values should be used. Caution: Permanent damage can result from forcing the output voltage higher than the input voltage. A protection diode from output to input should be used if this condition exists. Package Outline (5 Package - Steel) .295 (7 49) .265 (6 73) r-. ~ SEATIN~ 780 (1981)1 .76001(A1930) .057(145) .037 (0.94) PL"-A-N-E----yh-. Package TO-3 PO(MAX) = IJCA = IIcs Typ Max IIJC IIJC 1.8 2.5 .450 (11.43) .400 (10 16) t =* .043 (1.09) .038 (097) TJ(MAX) - TA IIJC + IJCA .161 (4.09) OIA .151 (3.84) 2 HOLES + liSA Solving for T J: TJ = TA + Po (OJC + IICA) Where: Junction Temperature TJ Ambient Temperature TA Power Dissipation Po Junction-to-case thermal resistance (JJC Case-to-ambient thermal resistance IICA Case-to-heat sink thermal resistance IIcs Heat sink-to-ambient thermal resistance liSA PIN 1 Notes All dimensions in inches bold and millimeters (parentheses) Pins are solder-dipped alloy 52 The device is designed to operated without external compensation components. However, the amount of external filtering of this voltage regulator depends upon the circuit layout. If in a specific application the regulator is more than four inches from the filter capacitor, a 1 ~F solid tantalum capacitor should be used at the input. A O. 1 ~F capacitor should be used at the output to reduce transients created by fast switching loads, as seen in the basic test circuit. These filter capacitors must be located as close to the regulator as possible. 4-26 SH1605 5-Amp, High-Efficiency Switching Regulator F=AIRCHILD A Schlumberger Company Hybrid Products Description The SH1605 is a hybrid switching regulator with high output current capabilities. It incorporates a temperature-compensated voltage reference, a dutycycle controllable oscillator, error amplifier, high current-high voltage output switch, and a power diode. The SH1605 can supply 5 A of regulated output current over a wide range of output voltage. • • • • • • • Connection Diagram 8-Pin TO-3 Type STEP-DOWN SWITCHING REGULATOR OUTPUT ADJUSTABLE FROM 3 TO 30 V 5 A OUTPUT CURRENT HIGH EFFICIENCY FREQUENCY UP TO 100 KHz UP TO 150 W OUTPUT POWER STANDARD 8-PIN, TO-3 PACKAGE • (Bottom View) Order Information Output Temperature Voltage Range 3 V To 30 V O°C to +70°C 3 V To 30 V -55°C to +150°C Block Diagram 51----------------------18 ~ 1 I7 I VOUT STEERING DIODE (ANODE) 1 I 1 I 1 11 GROUND I--_........~~..L.. CASE GROUND R1 TIMING CAPACITOR CT -..J._--I 1-_ _ _ _+--_-+--'VI1v-....,1,..-3 !~~~:IER INPUT R2 1 I I 1 1 12 L ______________________ -.J 4-27 g!~~~~~~~R NC~PINS6 Part Number SH1605SC SH1605SM SH1605 Absolute Maximum Ratings Vin - You! (Min) Input Voltage Output Current Operating Temperature (TJ) Operating Temperature (T A) SH1605SC SH1605SM Storage Temperature Internal Power Dissipation Duty Cycle Steering Diode Reverse Voltage Steering Diode Forward Current 5V 35 V Max 6A 150°C O°C to +70°C -55°C to +125°C -65°C to +150°C 20W 20% to 80% 60 V 6A Electrical Characteristics: T c = 25°C, TIN = 15 V, VOUT = 10 V unless otherwise specified. SH1605SC/SH1605SM Symbol Characteristics Conditions Min VOUT Output Voltage VIN 2: Vo + 5 V, 10 = 2 A 3.0 Vs Switch Saturation lOUT = 5.0 A, lOUT = 2.0 A VF Diode On Voltage lOUT = 5.0 A, lOUT = 2.0 A Vcc Supply Voltage IRo Diode Reverse Current VRO = 25 V 2.0 10 Quiescent Current lOUT = 0.2 A 30 /lA mA Typ Max Units 30.0 V 1.5 1.0 2.0 1.2 V V 2.2 1.6 2.8 2.0 V V 35 V 10 Reference and Oscillator Section XYREF Voltage on Pin 3 2.5 V l::,.V3/T V3 Temperature Coefficient 150 ppm/oC Xlc Charging Current-Pin 4 l::,.Vc 10 VIN=10V 20 VIN = 35 V 20 VIN = 10 V 150 VIN = 35 V 150 Voltage Swing-Pin 4 25 50 V 0.5 Discharging Current - Pin 4 /lA 70 225 250 /lA 350 Switching Characteristics (See Test Circuit) Symbol Characteristics Conditions tr Voltage Rise Time lOUT = 2.0 A lOUT = 5.0 A 700 1.8 ns tf Voltage Fall Time lOUT = 2.0 A lOUT = 5.0 A 700 900 ns ns ts Storage Time lOUT = 5.0 A 2.6 /lS td Delay Time lOUT = 2.0 A 2.5 /lS Min Typ Max Units /lS Thermal Characteristics Po Power Dissipation lOUT = 5.0 A VOUT = 10 V 16 W 1) Efficiency lOUT = 10 V VOUT = 5 A 75 % 8J-c Thermal Resistance 4.5 °C/W Notes 1. Typical is 30°C/W for natural convection cooling. 2. For heatsinking requirements see power derating curve. 3. VOUT refers to the output voltage range of a switching supply the output LC filter as shown in the typical application circuit. 4-28 SH1605 Switching Waveforms V3--------------.r---------------i CLOCK SIGNAL (PIN 4) V2-----r OV - - -- -+----...- - - - - I O F F - - - + i -------t~ v, -----------r (PIN a) WHERE V2-S0.2 V 4.0V" V3" 2.0V Switching Characteristics Test Circuit a 5 + + 3 1000 - SH1605 + ~F 15 V 4 .......--- ---- 1 7J: T'" ~ CLOCK SIGNAL Power Derating Curve 25 I/Jc=4.5"C/W CASE \. \ I'\. "JA~35·C/W o .0 30 FREE AIR eo r- I90 - 120 ~ '50 AMBIENT TEMPERATURE - DC 4-29 VL RL ~15"±10% SH1605 Design Equations Inductance = Ll = ( Vin(nom) - VOUT) X tON 611 . . POUT X 100 Efficiency (I)) = --"'---PIN tON Transistor DC Losses (PT) = louT X Vs -----'--tON + tOFF Where: . tOFF Diode DC Losses (Po) = louT X VF - - - tON + tOFF Drive Circuit Losses (DL) = V30lN02 X tON tON Switching Losses Transistor (Ps) = VIN X louT t, 2 (tON Transistor Duty Cycle = + tf + tOFF) ~::r: tON X (~_ 1\ VOUT ') tON tON Diode Duty Cycle = + tOFF + tOFF Delay Time = td = 2.5 /lS Typical Storage Time = ts = 2.6/ls Typical Nominal Input Voltage = VIN(NOM) Output Voltage = VOUT tOFF = 1 tON + tOFF Power Inductor (PL) = louT2 X RL (Winding Resistance) Efficiency (I)) = VOUT louT VOUT louT + PT + Po + DL Where: POUT = PIN lOUT Vs X 100 Output Power Dissipation Input Power Dissipation Output Current Darlington Switching Saturation Voltage Regulator "On" Time Regulator "Off" Time Steering Diode Forward Voltage Drop Input Voltage Regulator Switching Rise Time Regulator Switching Fall Time Output Voltage Inductor Winding Resistance [VOUT~::: R2)] - . Timing Capacitance (CT) Where: + Ps + PL VOUT SET RESISTANCE = Rs = Change in Inductor Current = 61 1 = 2 X louT(Min) Minimum Continuous Output Current = IOuT(Min) On Time = tON> (td + ts) tON is determined by the design and depends upon the desired frequency of operation under constant load conditions where frequency = 1/(ton + tOil). Off Time, toll, is determined by the ratio of input voltage and output voltage tON X Ie 6V e = --- Charging Current on Pin 4 = Ie = 25/lA Typical Voltage Swing on Pin 4 = 6 Ve = 0.5 V Typical Frequency = F = - - - - - - - CT 6Vc + 611 Ll Ie VOUT + VF Where: Steering Diode Forward Voltage Drop = VF = 2.2 V @ 5 A Typical (From Elect. Char.) = 1.6V @ 2 A Typical (From Elect. Char.) Minimum Output Capacitance 6it = COUT(MIN) (8 X F(MIN) X VRIPPLE(MAX)) [Rl + R2J = [2 X 103 VoUT - 2 X 103J VREF Where: Minimum Expected Frequency = F(MIN) F(MIN) = CT 6Vc Ie = 8 X 102 VOUT - 2 X 103 Typical + 6it (MAX) L VOUT + VF Maximum Change in Inductor Current Where: Internal Resistors = Rl = R2 =1X103 f! Reference Voltage On Pin 3 = VREF = 2.5 V Typical = AI I(MAX) = (VIN(MAX) - VOUT) X t ON Ll L> SH1605 Maximum Expected Input Voltage = VIN(MAX) Maximum Expected Ripple Voltage = VRIPPLE(MAX) Effective Series Resistance of = CONT = ESR VRIPPLE(MAX) L:. 11(MAX) Typical Application 300 ~H 8 5 3 C, 1000 SH1605 ; Rs ~F VII'; + 50V ;;;, -- 1 2 CT .O~~~ Rs = 1~F .-----~ - . - - - , 1- 1/=70% .875 (22.225) DlA MAX I I Load Reg. = 50 mV Line Reg. = 50 mV Ripple = 100 mV I .100~~ .085 (2.16) ~-IIf-r--'-----------'---'--+--'---'1 ---1 -'-...,.i-'-----,.,-"'TT-TT'"'TT"---'-f--'.......... t L.... VOUT VIN = 12-18V VOUT = 5V lOUT = 5 A (Max) lOUT = 1 A (Min) 'I .280-,----(7.,,) _ '_ .220 (5.59) t + + "' -I-- 2.5 Package Outline (S Package - Steel) Co 2000 ~F 50 V 7 ;;;, _ 03...:(c.. V",OU",Tc..-_ 2--.:... 5) 2_X_1_ .450 (11.43) .250 (6.35) -- CASE 4 Rs = VOUT Set Resistor L, ~ ~ ~ ~ ~ -II- SEATING PLANE .042 (1.07) CIA 8 PLACES .039 (0.99) TYP 1-_ _ _ '.197 (30.40) _ _ _.1 1.177 (29.90) .,--+--- ~~OCIRCLE .159 (4.04) CIA .154 (3.91) .188 (4.78) R MAX. 2 PLACES Notes All dimensions in inches bold and millimeters (parentheses) Pins are solder-dipped alloy 52 4-31 SH1605 Following is a partial list of sockets and heat dissipators for use with the SH1605. Fairchild assumes no responsibility for their quality or availability. a-Lead TO-3 Hardware Sockets Heat Sinks Mica Washers Robinson Nugent 0002011 Thermalloy 2266B Keystone 4858 (35°C/W) Azimuth 6028 (test IERC LAIC 3B4CB socket) IERC HP1-T03Augat 8112 - AG6 33CB (7°C/W) AAVID 5791B AAVID ENGINEERING 30 Cook Court Laconia, New Hampshire 03246 Azimuth Electronics 2377 S. EI Camino Real San Clemente, CA 92672 Augat P.O. Box 779 Attleboro, MA 02703 IERC 135 W. Magnolia Blvd. Burbank, CA 91502 Keystone Electronics Corp 49 Bleecker St. New York, N.Y. 10012 ROBINSON NUGENT INC. 800 E. 8th St. New Albany, IN 47150 Thermalloy P.O. Box 34829 Dallas, Texas 75234 4·32 F=AIRCHIL.O A Schlumberger Company 5-2 High Current Voltage Regulator Applications I=AIRCHILO A Schlumberger Company This application note is to assist the user in designing power supplies and on card regulation systems using Fairchild's family of series pass High Current Voltage Regulators. VOUT(Max) Selecting the Correct High Current Voltage Regulator The regulator selection guide (Table 1) provides a concise table of key regulator specifications by device number. Select the device that provides the desired output voltage and current, then proceed as follows. Maximum output voltage of regulator VOO(max) = 6. VL Maximum line voltage change VR(pk) = Maximum dropout voltage Peak ripple voltage = Also determine T A(max) = Maximum ambient temperature and select TJ < TJ(max) from the data sheet and see the application note titled "Thermal Considerations" for heat sink requirements. Determine the required input voltage (VIN). VIN(maX) = Design Precautions When designing and laying out a regulator circuit, follow these guidelines to save time, money and simplify design. > VIN > VOUT(max) + VOO(max) + 6. VL + VR(pk) where • VIN(max) = Maximum allowable input voltage VIN = Regulator input voltage under load Keep all ground leads as short as possible. Use ground conductors sufficiently large enough to handle rated currents to reduce unwanted voltage drops across leads, and to minimize heating effects and lead inductance. High Current Voltage Regulator Selection Guide ·S .. 0 :::J LL I: 41 U 41 SH323 0 U I: -.. . . - » 41 01 eu 0.- -:::J>e Q.eu ..5:!: Fixed Positive 40 78H05A Fixed Positive 78H12A 41 01 eu 0'- >2:."S~ .e-I: :::Jeu oa: I: I: I: 0 0 :::J eu :; :; 41 0.- -< :::J_ Q.>e "Seu O:!: I: ;( -:!: 1:_ 01 41 01 41 - ~~ 0... > o_ eu.- .- ii'; ...I~ 00 ""C 41..5~ 4I""C a:'ID .. 11141 41 .. :::J:::J o::.!! eu 0 U 41 'Qj' UI: 41_ a: ...1_ 41 01 0 eu a: . > :::J Thermal Resistance Max (OC/W) 0 41 01 eu .¥ Q. IJJC IJJA 2.5 38 U III n. 2-Pin 4.85 5.25 3 40 4.85 5.25 5 0.2 0.2 3 60 2.3 2.5 38 2-Pin TO-3 Fixed Positive 40 11.5 12.5 5 0.2 0.2 3.7 60 2.3 2.5 38 2-Pin TO-3 78HGA Adjustable Positive 40 5 24 5 0.2 0.2 3.4 60 2.3 2.5 38 4-Pin TO-3 79HG Adjustable Negative -40 -2.11 -5 24 0.4 0.7 -5 60 -2.2 2.5 38 4-Pin TO-3 78P05 Fixed Positive 40 4.85 5.25 0.2 1.0 3.4 60 2.5 1.8 38 2-Pin TO-3 0.2 0.2 3 60 2 TO-~ 10 5-3 High Current Voltage Regulator Applications • Use single-point grounding at the regulator common terminal whenever possible to prevent circulating currents or ground loops. • When using the adjustable multi-terminal regulators, especially at high output current levels, derive the feedback sense voltage from across the load rather than from across the regulator to improve circuit performance. • abuse and fault conditions that may be encountered occasionally. Continuous operation of the device under fault conditions such as a short or in a thermal shutdown mode is not a recommended procedure. Proper attention must be paid to the safe-area protection network when these regulators are operating with excessive input voltage or excessive input-output differential-voltage conditions. In addition to reducing the available output current with high input-output differential conditions, the safe-area protection network may, under certain conditions, cause the device to latch-up if the output is shorted to ground. This situation is aggravated as the input voltage, load current or the operating junction temperature is increased. This mode of operation does not damage the device but power (input voltage or load current) must be interrupted momentarily for the device to recover from the latched condition. High Current Voltage Regulators are particularly attractive because of the small number of external components required. It is good practice, however to use bypass capacitors at all times. Input bypass capacitors (1 ~F for positive positive regulators and 2~F for negative regulators) are especially critical if the regulator is located any appreciable distance from the power supply filter. Output bypass capacitors (0.1 ~F for positive regulators and 1.0 ~F for negative regulators) are also required to improve transient response. These bypass capacitors should be mylar, ceramic or tantalum with good high frequency characteristics. If more than one bypass capacitor source or more than one type is used, stability should be checked on each source or type. Stable operation with one capacitor from one vendor may not necessarily result in stable operation with a capacitor of the same type from a second vendor, since the characteristics of the capacitors may vary. Precautions must also be taken to avoid regulator operation beyond its absolute maximum ratings. Switching transients exceeding the maximum input voltage rating of a regulator, for instance, can destroy a regulator. These transients, which . occur especially if the regulator input voltage is switched instantaneously rather than ramped by the natural smoothing provided by the ac line and the filter capacitors, are usually hard to track and normally caused by lead inductance and fast switching currents. Good quality bypass capacitors that have low series resistance cause the inrush current to increase further, thereby causing a higher magnitude transient at the input of the regulator. In such cases, a lower quality and cheaper bypass capacitor may be the answer. Regulator output impedance is in the order of 100 ml1 or less and increases as a function of frequency above 10kHz due to the gain rolloff of the error amplifier. A tantalum electrolytic bypass capacitor connected to the regulator output will maintain low impedance for frequencies up to 1 MHz. A ceramic capacitor should be placed in parallel with the tantalum capacitor for driving fast switching loads to compensate for the rising impedance of the electrolytic capacitor above 1 MHz. If switching loads are distributed over a large area, additional ceramic bypass capacitors should be located at the loads. Very large-value output bypass capacitors should not be used unless adequate measures are taken to prevent the output from rising above the input, or to avoid discharging the bypass capacitor through the series-pass transistor of the regulator if the input is accidentally grounded. A reverse-biased diode connected from input to output is normally sufficient to achieve this protection. • Because of their output stage configurations, positive regulators source current and negative regulators sink current. These restrictions should be kept in mind and, under no circumstance, should a regulator output terminal be allowed to go more than a few volts higher than the regulated output of the regulator. The power should be turned off before removing or inserting a regulator into a test socket. However, if it is necessary to insert a regulator intoa "live" socket, care must be taken to ensure that the common terminal connection is made prior to, or simultaneously with, the input terminal connection. In the absence of the common terminal. connection, the output voltage of the regulator is 1 or 2 V below the input voltage. This type of fault condition can cause an excessive output voltage which may adversely affect the circuits supplied by the regulator. If the common terminal is quickly connected, the regulator can be dest~~yed. Also, damage to the regulator may result from the discharging of the bypass capacitor through the output and common terminals. Internal protection circuits are provided in all High Current Voltage Regulators to improve reliability and make these regulators immune to certain types of overloads. These on-chip components protect the regulators against short-circuit conditions (current limit), excessive input-output voltage differential conditions (safe-area limit) and excessive junction temperatures (thermal limit). The protection circuits protect the device against 5-4 High Current VOltage Hegulalor Applications • The thermal properties and limitations of voltage regulators are extremely important in circuit design. Whether or not a heat sink is required should be determined before the circuit is laid out. See the application note entitled "Thermal Considerations.' ' Adjustable regulators are ideal for applications that require non-standard output voltages. Output voltages are determined by the following equation: Applications A few of the most popular High Current Voltage Regulator Applications are illustrated in this section. These illustrations include both basic applications and some applications more exotic to extend the capabilities of the regulator. Where: R1 and R2 are set resistors as shown in Figures 2 and 3. VCONTROL = = 5 V(NOMINAL) for the /-lA78HGA -2.23 V(NOMINAL) for the /-lA79HG Output voltage can be set anywhere between + 5 V to +24 V for the /-lA78HGA and -2.11 V to -24 V for the /-lA79HG. A trimpot may also be substituted for R1 and R2 to allow for either full range adjustments or output voltage trimming. Basic High Current Voltage Regulator Configurations Figure 1 shows the basic connection diagram for fixed positive high current voltage regulators including the SH323, SH223, SH123, /-lA78H05, /-lA78H05A, /-lA78H12A and the /-lA78P05. The user may refer to Table 1 or the individual data sheets to determine which regulator satisfies his system needs. Fig. 1 Basic Fixed Positive High Current Voltage Regulator with Bypass Capacitors OUTt--,,-+VOUT +VIN-_~IN POSITIVE REGULATOR' COM CASE COMMON COMMON SINGLE POINT GROUND * Device Type SH323, SH223, SH123, /-lA78H05, /-lA78H05A, /-lA78H12A, or /-lA78P05 depending upon desired system parameters. See Table 1. Fig. 2 A Basic Positive Adjustable High Current Voltage Regulator r---------------~2 OUT 1--.._-_- +VOUT IN R, "A78HGA COM CONTROL 4 3 + COUT 0.1 "F R2 SOLID TANTALUM COMMON-----~~~~~==~-----COMMON SINGLE POINT GROUND Notes V OUT R1 + R2 = --R2 V CONTROL Nominal = 5 V Recommended R2 current VCONTROL ~ 1mA 5-5 High Current Voltage Regulator Applications Fig. 3 A Basic Negative Adjustable High Current Voltage Regulator OUT 4 IN t-......-....----....-+ • 0.001 ~A79HG -VOUT ~F COUT 1 COM CONTROL ~F + SOLID TANTALUM COMMON----~~~~====~---------COMMON SINGLE POINT GROUND 'May be necessary with long leads VOUT = ( R1+R2) R2 V CONTROL VCONTROL Nominal = -2.23 V Fig. 4 Parallel Operation of Regulators For Very High Current +VIN IN 2 OUT I - - - - _ - + V O U T ~A78H05 COM CASE IN ~A78H05 OUT 2 COM CASE IN ~A78H05 OUT 2 CIN l~F SOLID TANTALUM COMMON SINGLE POINT GROUND Parallel Regulators To obtain even higher output current, several regulators in parallel may be used. Regulation of the overall system can be improved if the individual devices are matched for output voltages as shown in Figure 4. If the outputs are not matched, it is likely that the output current will not be shared between the regulators and, as a result, some of the regulators will operate at or near the current limit while others are at their quiescent no-load levels. Excessive Input/Output Differential When a regulator is operating with a large inputoutput differential, the addition of a series resistor with the input extends the operating range of the device by sharing the power dissipation, see Figure 5. The value of the series resistor R1 must be low enough so that, under worst-case conditions, (lowest supply voltage, highest output voltage, and highest load) the device remains in regulation. R1 can be calculated as follows. 5-6 High Current Voltage Regulator Applications R1 = Fig. 5 Reducing Power Dissipation in a Regulator with Dropping Resistor R1 VS(min) - VOUT(Max) - VDD(maX) louT(max) + IO(max) where RI Vs(min) is the minimum supply voltage VDD(max) is the maximum dropout voltage IO(max) is the maximum quiescent current VOUT(maX) is the maximum output voltage = = Line Regulation (Max) ------'~--~~-- + lo(max)] R1 120 mV X 6.6 V 25 V - 16 V 88 mV From PD(max) = (35 - 11.5) X 3 70.5 W (without R1). = To PD(max) = (35 - 3.3 X 3 - 11.5) X 3 = 40.8 W (with R1) Note that the power dissipation is shared between the regulator and R1. Load regulation at constant Vs = load regulation at constant VIN + line regulation Although bypass capacitors are not shown in Figure 5, it is recommended that they be incorporated in the design as illustrated in Figure 1. Example: Assume a supply range of 25 to 35 V used with a J.LA78HG12A regulator delivering an output current of 1 to 3 Amps. Input Voltage> VIN(maX) When a regulator is used with supply voltages greater than the rated regulator maximum input voltage, the circuit shown in Figure 6 can be used. This circuit essentially provides a constant voltage to the regulator with supply voltage variations. The choice of Zener diode voltage is dictated by VIN(min) of the regulator and VSE(max) of Q1. From the data sheet: VOUT(min) = 11.5 V VOUT(max) = 12.5 V = = The inclusion of the 33 Q reduces the maximum power dissipation of the regulator as shown below. The load regulation can therefore be calculated as follows. IO(max) X 6 VIN = The effect is 88 mV additional change at the output terminal. For load regulation, assuming constant supply voltage, the combined effects of the change at the input due to the voltage change across R1 must be taken into consideration. In this configuration, as the load is increased, the regulator input voltage decreases and the net result, in most cases, is a slight degradation in the performance of the regulator since these two effects are additive. = lOUT 6 VIN (For Line Regulation Test) For a constant load, the regulator input voltage varies by the same amount as the supply voltage and consequently the line regulation of the device remains essentially the same. VDD(maX) COM The effect of the 6.6 V change at the regulator input under worst case conditions can be determined from a ratio of data sheet parameters: [VIN(max) - VOUT(min)]loUT(max) VS(max) - [louT(max) 2. OUT ~ CASE1 • IQ where VIN(maX) REGULATOR IN Vs Maximum regulator dissipation, however, occurs with highest supply voltage and highest load current. PD(max) VIN 2.5 V 10 mA R1 = 25 - 12.5 - 2.5 = 3.3Q 3 + .01 With this value of R1 and a load varying from 1 to 3A, the input voltage to the regulator varies, 6 VIN = 610uT R1 = 6.6V 5-7 • High Current Voltage Regulator Applications Fig 6 Regulator Input Circuit for Input Voltage Source Greater than VIN(max) REGULATOR +vs OUT IN +VOUT i----r---'" + COUT 0.1 ~F COMMON------~==::::==~~~----------COMMON Fig. 7 High Output Voltage Regulator, No Short-Circuit Protection IN +VIN POSITIVE REGULATOR C,N CASE OUT 2 +VOUT +COUT 0.1 R,0PTIONAL SEE TEXT ~F COMMON--~----------~--------------COMMON Fig 8 High Output Voltage Regulator with Short-Circuit Protection Q2 2 OUTt-_---.._-+VOUT COMMON COMMON voltage exceeds the maximum input voltage rating of the regulator. Figure 8 can be used to take advantage of the protective features of the regulator. Here too, the regulator common terminal operates on the pedestal established by Zener diode D1. Zener diode D2 and the Darlington configuration of 01, 02 reduces the regulator input voltage to a safe value. The Darlington configuration prevents loading of Zener diode D2, and thus maintains a high level of regulation. Diode D3 protects the regulator against accidental shorts by clamping the common terminal of the regulator to a diode drop above the shorted output. High Output Voltages Figure 7 shows a simple circuit that can be used to obtain an output voltage greater than the standard fixed voltages available. The quiescent current biases Zener diode D1 and the regulator common terminal rides on the pedestal established by D1. If the Zener must be operated at currents greater than the quiescent current level of the regulator, then R1 can be used to increase the Zener current. If, on the other hand, lower Zener current is satisfactory, R1 can be placed in parallel with D1 to shunt some of the current. Caution: this circuit configuration cannot utilize the thermal shutdown or short-circuit protection features of the regulator if the input 5-8 High Current Voltage Regulator Applications The input voltage VIN must be high enough to accommodate the dropout voltage at the low end, but must not exceed the maximum input voltage rating at the high end. Remote Shutdown Electronic shutdown is used in some applications where, under certain conditions, the removal of power from the load is desired. The 3-terminal regulator circuit of Figure 9 has a remote shutdown feature. Under normal conditions, 02 is on and provides the base current of 01 . Positive and Negative Adjustable Regulators The concepts used above for positive fixed regulators can easily be extended to the JlA78HGA, positive adjustable regulator, by simply including the R1, R2 resistor network shown in Figure 2. 01 acts as a switch and is either in saturation, when the signal to the base of 02 is high, or is off when the signal to the base of 02 is low. It must have a current rating equal to the load current. Turn-off time is dependent on C2 and the load current; the higher the load current, the faster the turn-off time. Also, since negative voltage regulators are complements of the positive voltage regulators, almost all the positive regulator applications can be converted into negative versions by appropriate changes in the polarity of the input voltages. If external active components such as series-pass transistors are used, they should be the complements of those used in the positive-regulator application, i.e., npn transistors replaced by pnp and vice versa. Finally, these concepts can be extended to the JlA79HG, negative adjustable regulator, by simply including the R1, R2 Resistor Network shown in Figure 3. Constant Current Regulator Any regulator can be used as a constant-current regulator as shown in Figure 10. The current lOUT which dictates the regulator type to be used is determined by this equation. lOUT = VOUT -- R1 + 10 where VOUT is the regulator output voltage and 10 is the quiescent current. Fig. 9 Remote Shutdown +VIN _ _ _--"' 0, POSITIVE 2 REGULATOR OUT IN +VOUT COM CASE CIN +COUT +1"F O.1"F SOLID TANTALUM COMMON-----~-~-~-~----~COMMON Fig. 10 Constant Current Regulator (Positive Output) ~J... IN POSITIVE OUT REGULATOR COM ::; CIN SOLID TANTAL U'; 1"F CASEl L 2 t VOUT Q, _ ~ ;;; COUT ::;O.1"F R, ,.: COMMON 5-9 lOUT ~ High Current Voltage Regulator Applications Dual Regulators Dual regulators, or dual power supplies, are normally used for applications requiring two output voltages of opposite polarities that do not necessarily have equal magnitudes, for example, +12 V, -5 V. However, the word dual can also imply two supplies of the same polarity but of different magnitudes, such as +5 V, + 12 V. With dual tracking, not only are the output voltages of different polarities, but one output voltage always follows the other one, i.e., an increase in the positive voltage results in a decrease in the negative output voltage. Dual Supplies The simplest dual-polarity high current supply can be obtained by using a positive and a negative adjustable regulator with a center-tapped transformer as shown in Figure 11. The same type of dual supply can be achieved with two positive (or two negative) adjustable regulators if a transformer with two isolated windings is used as shown in Figure 12. The reverse-biased diodes connected across the outputs of the dual regulator circuits are not necessary if the loads are referenced to ground. If the loads are tied between the two outputs, however, a latch-up may occur at the instant power is turned on, especially if one regulator input voltage rises faster than the second one. The diodes, that ensure proper start-up of the regulators by preventing a parasitic action from taking place when power is turned on, should have a current rating equal to half the load current. Fig. 11 Dual Supply using a Center Tapped Transformer with a Positive and a Negative Adjustable High Current Voltage Regulator r---- --.J.- CASE GROUND R1 TIMING CAPACITOR C, --''-----I I I I , 1 - - - - - + -_ _""""'V\tv--r'-3 !~~~~'ER INPUT R2 , ______________________ L 4-27 '2 g!~~~~~~~R ~ ~-~6 Part Number SH1605SC SH1605SM High Current Voltage Regulator Applications Fig. 13 A Dual Positive Supply with a +5 Vand +10 V Output 2 +VIN-.....- - - - - - - - 1 I N "A78HOS A OUTt-<~........--VOUT 10 V COM 0, CASE IN "A78HOS B ...-----'-----VOUT SV OUT t-_-~ COMMON-~----~----.....~--4--------------COMMON Fig. 14 A Dual Polarity Supply from a Single Transformer Winding SECONDARY WINDING IN POSITIVE REGULATOR OUT A lOUT + A +VOUT COM RL+ CASE IN POSITIVE IOUTB REGULATOR OUT B 2 COMMON RL -CASE Figure 13 shows a single positive-polarity dual supply with + 5 and + 10 V output. It uses two 5 V !LA?8H05 regulators operating from a single positive voltage source. The + 10 V output is achieved by connecting the common terminal of the top regulator A to the output of the bottom regulator B. Diode 01 ensures proper start-up of the top regulator and prevents a latch-up that may occur under a heavy load condition on regulator A. Resistor R1 provides a path for the quiescent current of regulator A and can be eliminated if regulator B has a minimum load current greater than the quiescent current of regulator A. -VOUT adjustable regulators. Because of the internal feedback of the 4-terminal regulators, a constant voltage is developed across the resistor string R1, R2 and R3. Variations at one of the output nodes are reflected at the control nodes causing corresponding variations at the opposite output node. Note that tracking between the two outputs is not one to one but rather depends on the absolute value of the two references and the feedback resistors R1, R2, and R3. The output voltages are determined by VOUT+ = VREF+ R1 +(VREF+ - VREF-) R2 The concept of Figure 13 can be used to achieve a dual-polarity output from a floating single supply as shown in Figure 14. This circuit is restricted in that RL + > RL -, since all of the current provided by the positive regulator A must flow through RL -. VOUT- = R4 VREF - - (VREF+ - VREF-) R3 Tracking between the outputs can be improved by adding a !LA? 41 and modifying the circuit as shown in Figure 16. This circuit yields an adjustable true dualtracking regulator with internal short-circuit protection, safe-area limiting, thermal overload protection, and is capable of a 5 A maximum output current. The outputs of the regulators are independently adjustable by potentiometers R1 and Adjustble Dual Tracking Regulators For applications requiring adjustble tracking outputs, the circuit of Figure 15 can be used. Tracking is accomplished by connecting a common resistor between the control terminals of the two 4-terminal 5-11 High Current Voltage Regulator Applications R2. With the component values shown, the output voltage of the positive JlA78HGA can be varied from 5 to 24 V, and the negative JlA79H6 can be varied from -2.11 to -24 V. follows: any change in the positive regulator output causes an opposite change on the common terminals and also on the negative regulator output. For example, a decrease in the positive regulator output voltage causes a like change in the amplitude of the negative regulator output. Since each regulator has a reference, no slaving exists between the outputs and, as a result, tracking is true and independent of polarity. This circuit has a positive and a negative regulator and an operational amplifier used as a comparator. Tracking is accomplished by connecting the two regulator common terminals to the output of the JlA 741 that provides a potential on which the common terminals of the regulators float. The summed regulator outputs, Vo+ and Vo-, are then compared to the power supply common. Proper care must be taken to insure that the maximum supply voltage rating of the JlA741 is not exceeded when the regulators are operating with high input voltage sources. The positive and negative regulator outputs track as Fig. 15 Adjustble Dual Tracking Regulator OUT L~ IN "A78HGAI "A79HG + "F +VOUT + r"0.1 "F Fl1 caNT COM 1 COMMON COMMON +~hF + R2 COM CONT 1 "FJ,. ~ IN "A78HGA/"A79HG OUT 'f~~ R3 Fig. 16 Independently Adjustable True Dual Tracking Power Supply OUT 1 IN COM ;;; L~ "A78HGA + O.33"F +VOUT R1 25k CONT 3 4 +", O.l"F'"r- R3 5k COMMON ~'7 "A741 3V 6 R4 4 2.2k *+ l"F 1 + 2"F 4 -v IN IN CONT~ "A79HG ++ OUT .1 3 r: R2 i> 25k VOUT COMMON -'- 5-12 High Current Voltage Regulator Applications Miscellaneous Applications This section consists of a set of illustrations showing a variety of additional high current voltage regulator applications. Fig. 17 Negative Output Voltage Circuit ......------~--- r-----<~----- ~II ~OUTPUT + POSITIVE REGULATOR ':" Fig. 18 Programmable Supply +10V +35 V TIMER COUNTER I 20 k 16 1 3~ 15 2 ;6 k 20 k 14 3 8k 4 4k 5 2k 13 1 "F 2240 0.1"F + 10 V F 1 10 k 11 OUT 0---- 10 k 10 ....-+---4>--W\r-.......-t "A78HGA --- Output Waveform 30V------/ / ~/64STEPS 5vI OV----------_______ 5-13 VOUT IN 2.5 k COMMON CONTROL I---' " 0.1 "F • High Current Voltage Regulator Applications Fig. 19 Motor Speed Control OUT ~A78HGA 25k IN +V'N 12 TO 20 V DC MOTOR R, 0.1 ~F I ........ VOUT = V RI ( 1 + :~ ) + IQR2 5-14 R2 OUTPUT High Current Voltage Regulator Applications Fig. 22 Signal Driver/Modulator 1 2 ~A78H05 0.1 3 OUTPUT 1 "F 47 SIGNAL INPUT n 1 l. Fig. 23 High-Current ECl Regulator Using ILA79HG Out 5.2 "A79HG Rl 2.7 K 1% In Unr Inpu r 33 on Control 2.0 '~"r' ~F Common V VOUT ~ 1 ~F R2 2 K 1% _ .... 'Solid Tantalum Close to Device Fig. 24 Operational Amplifier Supply (± 15 V @ 5 A) +15 V OUTPUT 10 kn +20V INPUT + 1 ~F IN4001 OR EQUIVALENT 0.1 "F 3 7.5 kll GND GND 2.0~F 2 -20V INPUT 4 + 1 ~F ~A79HG IN4001 OR EQUIVALENT 12 k!l 3 -15 V OUTPUT 5·15 Understanding the Switching Regulator FAIRCHILO A Schlumberger Company keep components small and switching inaudible. Filter F serves as an averaging filter, converting the pulse from S into a dc voltage. Assuming no losses, the power in equals the power out: A basic switching regulator is composed of four major components: a voltage source Ein, a switch S1, a pulse generator Ep, and a filter F1. The block diagram in Figure 1 shows the interconnection between these elements. The voltage source may be any dc supply needing conversion and/or regulation - a battery, an unregulated rectified and filtered supply, or even a regulated voltage to be converted into some other required voltage. The requirements for a voltage source are: • It must be capable of supplying the required output power plus the losses associated with the switching regulator. Pin = Pout • The input voltage must be sufficiently high to overcome any IR drops and meet the minimum requirements of the system. • The input voltage must be large enough to supply sufficient dynamic range for line and load variations. • This switching mechanism allows a conversion similar to transformers, thus the switching regulator has often been referred to as a dc transformer. The relationship of input and output voltage is a function of duty cycle. The duty cycle is the ratio of the ontime (ton) to the period (tp = ton + toft = 1/f). Thus, the duty cycle (J = ton ton + toft , Eout = (J Ein = ( t o n ) Ein ton + toft With (ton + tott = tp) constant, the output voltage Eout is directly proportional to the on-time ton. Thus, varying ton varies the output voltage (i.e., pulse width modulation). With ton held constant, the output VOltage, Eout, is inversely proportional to the period, tp = ton + toft, or directly proportional to the frequency, f = 1/(ton + toft). In a modern computer power supply, the input voltage may be required to store energy for a specified amount of time during brownouts or power failures. These techniques allow efficient generation of low voltages from high voltages in a stepdown regulator. Operating from voltages much too high for linear conversion affords a wide dynamic range and high energy input storage for brownouts and missing cycles. Fig. 1 Basic Switching Regulator The Filter The filter or integrating network is of major importance in the proper design for the switching regulator. The filter basically has three forms: RC filter RL filter RLC filter While all these filters are used in switching regulators, the RLC filter is most often used in series switching regulators. A brief analysis of the RC and RL filters gives the foresight needed for understanding the RLC filter design. SwitchS is typically a transistor or thyristor connected as a power switch. The switch is inherently efficient because it is operated in the saturated mode. The pulse generator alternately turns S on and off. The pulse is generally an asymmetrical square wave varying in either frequency (frequency modulation) or pulse width (pulse with modulation). Theoretical analysis and formulas generally apply to both frequency or pulse width modulated systems. The frequency f1 of the pulse generator is usually in the tens of kilohertz to 5-16 Understanding the Switching Regulator The RC Filter A simple switching regulator employing an RC filter is shown in Figure 2. When a switch 01 closes, the instantaneous current in capacitor C1 is very large and limited only by the series resistance Rs and the ESR* of the capacitor. This instantaneous current can be found from Kirchhoff's Voltage Law, using Laplace transforms. The resultant formula is the familiar equation for finding the current in an RC circuit. It can easily be seen that at t = 0 +, the current is limited by R only. When switch 01 is open, the voltage across C1 starts to decay in accordance with the formula: (See Figure 2) E S - ~ - IsR + _15_; CS Is ( R + R = Rs + ESR* In order to maintain the voltage on C1 (Le., Eout) relatively constant, it is necessary to make the charge time constant much shorter than the load time constant ds ) *ESR = Effective Series Resistance = Is E = ~ Q As R becomes smaller, the averaged square wave approaches a dc source. However, as R becomes smaller, the peak current Ie becomes larger. These peak currents are very high and impractical to switch reliably. As R is increased to limit the current, it becomes noticeably lossey and dissipates excessive power. -1 a= RC - - - eat The Rl Filter A simple switching regulator employing an RL filter is shown in Figure 3. As switch 01 closes, the voltage across the inductor is the full power supply voltage E. The current supplied to the load at t = 0+ is approximately equal to zero and exponentially s-a -t Is = E R -e RC Fig. 3 Simple Switching Regulator with Rl Filter RS Rs + Ee EL ~ E (INDUCTOR) TIME 5-17 Understanding the Switching Regulator The RLC Filter Combining the RC and RL filters gives all of the advantages of both, with few of the disadvantages. Figure 4 depicts the RLC filter in the simple switching regulator. The inductor L 1 is used to limit the peak currents associated with the charging of capacitor C1. This current will be highest during the initial turnon of the power supply with all initial conditions set to zero. This circuit is shown in Figure 5. The peak current then is again derived by Laplace transforms from Kirchhoff's Voltage Law. increases as shown in the curve in Figure 3. In a similar fashion, the instantaneous current can be found using Laplace transforms and Kirchhoff's Voltage Law: E S = IsR + IsLS; . . . RE[1 Yielding: IL = R = Rs + Rinductor ~ e ~RtJ -L~ The resultant formula is the familiar equation for finding the current in an RL circuit. It can easily be seen that at t = 0+, the current is zero. Thus the time constant L/R must be smaller than the load time constant to average the square wave into a dc source. L1 L1 R RL - E S 1 = IsRs + IsLS + IsRL l i CS E S -«Is While the inductor does overcome the large peak current phenomenon of the RC filter, there are three important disadvantages associated with the RL filter. • Since the current cannot change instantaneously in an inductor, a sudden change in the load (RL) will cause an abrupt change in the output voltage. This phenomenon is the limiting factor that determines the transient response of a switching regulator. • The energy stored in an inductor is determined from the equation e= 1 2 [ R~S+1 RLLCS2 + (Rs R L C + L) S + Rs + RL U2. Is Since the energy changes with the square of the current, the inductor must be very large to provide constant current flow when the load current is small. • [~Jx The disruption of current in 01 (shutting 01 off) causes the magnetic field associated with L 1 to collapse and induce a potential in accordance with Lenz' law: Resulting in the form: This negative voltage places a very large voltage across transistor switch 01 and will probably result in its destruction. VCE(Off) = Ein f(s) = + IeL I 5-18 S (S S+d a) (S ~ ~ b) ] Understanding the Switching Regulator Yielding the general equation: I = Ae at + Be bt + K A= a+d a(a - b) B= b+d b(b - a) K= d ab diode D1 steers the current developed by the collapsing magnetic field and charges capacitor C1 during the off-time; D1 also acts as a clamp and limits the negative potential to one diode drop. Diode D1 is called a steering diode, com mutating diode or free wheeling diode. This circuit not only protects switch 01, but also uses the energy stored in the magnetic field to charge capacitor C1 ; thus, Ll2 = CE2. Optimization of the RLC filter requires examination of the current loop equation for the RLC filter. Figure 5 depicts the RLC circuit used for the analysis. This formula is used in the section analyzing a typical switching regulator. A typical curve for Is is shown in Figure 6. Under light load conditions, the capacitor C1 supplies the necessary current to the load in accordance with the equation e = ~ CE2. 2 -R Under 8, =--a-+ heavy load conditions, the energy stored in L1 supplies the current in accordance with the equation e= -R 82 = - - + 2L ~ Ll2. The energy stored in the magnetic field 2 that resulted in the negative induced voltage can now be applied as an advantage. As shown in Figure 4, Fig. 4 Basic Switching Regulator with RLC Filter 0, V A =_0 2L 1= Aes,t - Aes2t; V V R2 1 4L2 - LC R2 4L2 1 LC Fig. 6 Turn-On Peak Current c, Fig. 5 Equivalent Circuit (Turn-On) Fig. 7 Frequency Response Curve of an RLC Filter RL A: CRITICALLY DAMPED B: OVERDAMPED C C: UNDERDAMPED TIME 5-19 Understanding the Switching Regulator The waveforms for the dual circuit are the same as those in Figure 7. Thus, to insure that for both on and off conditions of switch 01, both circuits are slightly overdamped. Examining the roots of the equation shows that three special conditions exist: * Underdamped case: R2 1 4L2 < LC ; W<0.5R V-f< The solution is complex and is exhibited as an oscillatory condition. This condition is undesirable due to the associated losses, i.e., energy in the ringing, and the RFI produced. See Figure 7. W>0.5R In the off-condition, RL is variable, with the worst case occurring during light loads. This can be alleviated with a minimum RL or a snubber network as used in the on-condition. As aforementioned, meeting these inequalities enhances both the RFI characteristics and the possibility of parasitic oscillations. The solution for these roots is real. This condition is also undesirable in the extreme case due to its associated losses. See Figure 7. Critically damped case: 4L2 LC 0.5 RL (off-condition) In the on-condition, Rs is a fixed quantity and should be made small to minimize IR losses. Thus, a snubber network may be required to dampen any oscillations associated with a small Rs. Overdamped case: ~ __1_ 0.5 Rs (on-condition) ~CL "\ V i- Overshoot and Undershoot When the load is abruptly changed (i.e., load current), the output voltage changes accordingly. This is called overshoot for decreasing loads and undershoot for increasing loads. Expressions for overshoot and undershoot can be derived from the two equations: 0.5 R = The equation has a real solution and is the most desirable case since losses are at a minimum. However, since this is not a practical case state, the circuit is operated in a slightly overdamped condition. See Figure 7. di -eL = L dt During the off-condition of switch 01, the circuit becomes the dual of Figure 5. This too has three similar conditions: Underd ampe d case V-f - L where eL = Ein - Eout for increasing loads eL = Eout for decreasing loads di = ic = change in load current = 61 t = transient time dv = overshoot/undershoot voltage 0.5 >R VIc -t <0.5R Overdamped case "\ ·· II y d amped case C ntlca V-f - L . IC= 0.5 = - C - dv . d' IC= I dt ' dt=C R Ldi eL = ~ di C ~ di Ldi 2 dv=-C eL 6 Eout = L 61 2 f . Ioa d s or'increasing C (Ein - Eout) L 61 2 6Eout=-C Eout 5-20 for decreasing loads Understanding the Switching Regulator Transient Response is directly proportional to the inductance. Item 5, the circuit overshoot The transient response, as mentioned earlier, is limited by the size of inductor L1. This transient response time tR is the time necessary before the system can compensate for an abrupt change in the load, assuming zero loop response. Transient response time can be found from the equation: [ .6Eout = L .61 2 C (Ein - Eout) J is directly proportional to the inductance. Item 6, the size and cost are directly effected by the inductance as well as a host of other factors. di -eL= L dt 2L .61 Item 7, the effect of the inductor on radiated electrical and magnetic noise is a function of geometry, frequency, size and cost. It becomes apparent that selecting the inductor requires careful consideration of the aforementioned tradeoffs. Applications of these tradeoffs are considered in the analysis of a typical switching regulator. for increasing loads Ein - Eout for decreasing loads The Inductor The inductor is perhaps the least understood of the switching regulator components and yet one of the most important. There are seven major areas with tradeoffs to be considered. 2. Peak current limiting in 01 Inductor design has many philosophies associated with it. Size constraints are radiated electrical and magnetic fields may dictate a powder toroid or pot core; however, in most applications (computer and peripherals), the decision is left to the design philosophy. The three most common techniques employed in the industry are: 3. Output ripple • Powdered permalloy toroids 4. Transient response • Ferrite EI, U and toroid cores 5. Overshoot • Silicon steel EI butt stacks 6. Size and cost limits The first technique, the powdered permalloy toroid, yields perhaps the most stable and predictable inductor. Powdered permalloy toroids have low leakage inductance, high permeability and low core losses. The major disadvantage is the cost of manufacturing and mounting toroid inductors. 1. Energy storage for the regulator 7. Radiated electric and magnetic fields As inductance is increased, items 1 through 3 are enhanced. Item 1, the energy (e = ~ L 12) is 2 The ferrite EI, U and toroid cores exhibit low losses. The ferrite toroid has low leakage inductance but is as expensive to manufacture as its powdered permalloy counterpart. All ferrite cores have low permeability, poor high temperature performance and the expense in mounting. The silicon steel EI butt stack offers one of the best tradeoffs in low voltage switching regulators. The silicon steel laminations exhibit high permeability, high flux densities, ease of construction and mounting. Core losses, while higher than the powdered permalloy and ferrite cores, are usually insignificant at low voltage levels. The silicon steel lamination is a common material in most magnetic houses and often can be found on the shelf. directly proportional to the inductance. Item 2, the peak current, is inversely proportional to the inductance. Item 3, the ripple voltage [ ERIPPLE -- Ein - Eoutl 41!'2f2LC J is also inversely proportional to the inductance. However, as the increase in inductance enhances operation of items 1 through 3, it is detrimental to items 4 through 6. In item 4, the transient response [ tR = 2.61 Ein - Eout J 5-21 Understanding the Switching Regulator The magnetizing from Ampere's Law is found from the equation: Inductor Design Combining Faraday's Law and Lenz' Law yields: E N = ~ X 10-8 ~ L = dt Hdc dt = 0.4 Nldc The inductance is found from the equation: L N d4>x 10- 8 = 3.19N2Ac X 10- 8 dt = .f L Ig 6J.L d' _I = LI BAeN X 10- 8 = Incremental permeability can be found from manufacturers' data sheets as shown in Figure 8. Linearity of the inductor can be enhanced by making Ig large. dt Multiplying both sides by ~ L 12 = ~ yields: 2 Units: Ac = cross sectional area (in.2) Bac = ac flux (Gauss) Bdc = dc flux (Gauss) f = frequency (Hz) Hdc = magnetizing force (Oersteds) Ie = mean length of core (in.) Ig = gap (in.) N = number of turns 6J.L = incremental permeability B Ae NI x 10- 8 2 2 which is the energy stored in the inductor. Integrating Faraday's Law E flux in the core. = N de/> yields the formula for ac dt 3.49 E x 106 G auss f Ac N Bac There are several off-the-shelf inductors manufactured by Sprague called Soft Inductors. The Soft Inductor is designed specifically for switching regulators, with a special variable reluctance gap. The dc flux is found from the equation: Bdc = +..!£ 0.6 Nldc Gauss Ig Figure 8 10,000 130 sooo o.~ ,/ 100 . r---- H, 0 >>10 20 50 100 200 .......... 1"\ 0 "-["'\ I)" 2 r\ 0 5 \ ¥' . 500 1000 2000 5000 10,000 o ~ ~ ~ ~ ~ m [\ \ _ ~ PERCENT OF RATED DIRECT CURRENT BMAX(GAUSSJ a. Incremental permeability curve for AISI grade M- b. Effect of dcin a typical filter choke. Inductance drops linearly until rated dc is flowing through coil, then drops rapidly as core saturates. The linear portion of the curve has less slope for inductors that have larger air gaps. 22 laminations where Ho is the dc magnetizing force in core. 5-22 Understanding the Switching Regulator The Output Capacitor Selection of the output capacitor also requires care. Consideration must be given to both the ESL and the ESR. Very often the ESR contributes more to ripple and noise than its reactance does. Desirable characteristics can be achieved by carefully paralleling three or four different types of capacitors such as tantalum, electrolytic and ceramic capacitors. Capacitors especially developed for switching regulators are now available in a multitude of ranges, sizes and types, with low ESR and low ESL at the switching frequencies. A curve of the 4terminal UFT capacitor (manufactured by Cornell Dubilier) compared to a conventional electrolytic is shown in Figure 9. The curve plots impedance versus frequency. The UFT capacitor remains quite flat beyond 1 MHz. The UPT capacitor, also manufactured by Cornell Dubilier, is designed for switching regulators and gives one of the best performance/cost tradeoffs available. A simplified equivalent circuit is shown in Figure 10. = Z3 = Rdo Z2//Z3 ,= + Ij(XESL - Xc) (Rs + rs + ESR) + j(X2 + XESL ESR ------~---~--- ERIPPLE E [ (Rs In ERIPPLE Xc) = = ESR + j(XESL - Xc) + rs + ESR) + j(Xc + XESL - ] Xc) 'Y Ein The formula: C1 = Ein - Eou! 4 7r 2f2LERIPPLE is a good approximation for finding the minimum capacitance; however, the preceding formula must be used to accurately determine the ripple voltage. ~ is w ~ jXc ,= ------Z1 + Z2//Z3 Figure 9 Closed Loop In order for the switching regulator to maintain an output voltage relatively constant, some feedback mechanism must be employed. Figure 11 shows a typical feedback system. 0.1 i • K3 represents the power switch, filter and all the associated losses. • K2 represents the transfer function for the pulse generator. FREQUENCY - HERTZ Fig. 10 Simplified Equivalent Circuit of RLC Filter ,------, Rs Z2 + rs + jXL ESR + jXESL - Z1 = Rs • K1 is the open loop gain of the error amplifier. I • {3 is the attenuation factor usually determined by a simple voltage divider. I L _ _ _ _ _ _ .-.l • ~ is a summing network that produces an error voltage 6e from the difference between the reference voltage EREF and the feedback voltage I Rs Efb. ,- - - l I I n EIN I I ESL I I I I L_ Xc II ESR The total loop gain will determine the percentage regulation of the switching regulator. I I I I I I A= RL 6 Eou! = % regulation 6E The error amplifier gain then is defined by: I K1 __ J = A-1 {3K2, K3 While this model is an approximation, it yields relatively close results. 5-23 Understanding the Switching Regulator Fig. 11 Typical Switching Regulator Feedback Loop EREF ,-----, r-----l I I I I I I ---If---I I I I I L __ SUMMING I L _____ ---1 NETWORK PULSE WIDTH MODULATOR ERROR AMPLIFIER I I I I I I I I I I I L _____ J I SWITCH, FILTER & CIRCUIT LOSSES r-------, I I I I ~Efb tJ = -_._-~Eo ~Eo I I I L _______ J FEEDBACK NETWORK Therefore, a switching regulator maintains a constant output voltage against variations in input by appropriately modifying the system duty cycle. The basic switching regulator operates as follows. Control transistor S1 switches on when first energized, thereby applying a voltage approximately equal to the input across L 1 and Co. This causes current h to increase linearly with time, supplying current to the load while storing energy in L 1. Diode D1 insures that, when S1 switches off, current continues to flow to the load thereby achieving a continuous load-current flow. A Switching Regulator Using the SH1605 The SH 1605 is a hybrid integrated circuit, designed specifically to be used as a major building block in high-current, step-down switching regulator systems. It contains a temperature-compensated voltage reference, comparator, oscillator, highcurrent Darlington and high-power steering diode. This device is capable of supplying up to 5 A continuous current; its package dissipation capability is 20 W maximum. This circuit provides excellent performance, with efficiencies up to 85%, for applications requiring high power densities and large operating currents. At equilibrium, the average current through L1 is equal to the load current. The rate of current change Switching Regulator Theory Figure 12 shows the basic switching regulator configuration. This circuit provides an output voltage VOUT, related to the input voltage V IN by the duty cycle of switch S1. Thus: VOtJT = VIN ( ton ton ) + toll Fig. 12 Basic Switching Regulator L l' VOUT '----+__ TO CONTROL CIRCUITRY COUT 5-24 I Understanding the Switching Regulator SH1605 Theory of Operation The SH1605 simplified block diagram is shown in Figure 14. Circuit operation is as follows. When power is first applied, the output voltage VOUT is low, thus forcing the comparator output into a HIGH state. As a result, the oscillator freely toggles the output switch on and off at a rate determined by the charge and discharge rate of the timing capacitor CT. This is a temporary condition that continues until VOUT has exceeded the reference voltage level times the factor set by Rs, R2 and LRs. The output voltage can be expressed as follows. through the inductor, 61" during the on and off period is defined by Equations 1 and 2 below. 6h (VIN = ~ 1VOUT ) ton (1 ) (2) Since, in a conventional switching regulator the excursions 1' (on) and 1,(011) are equal, Equations 1 and 2 can be written as follows: ~-1 = toft VOUT ton V - V (Rs + R2 + Rs) OUT - REF (R1 + R2) (3) Since the value of R1 and R2 (1 krl each) inside the SH1605 is established, Rs can be determined as follows. Equation 3 shows the natural tendency for the on-tooff time ratios to remain proportional to the inputoutput voltage differential. Voltage regulation can be achieved when this information is properly fed back to the switch. Rs = (2 X 103 ) (VOUT - 2.5) for Rs in ohms 2.5 13 Typical Switching Regulator Waveforms VOUT + V01 L1 OA I I I I I I I I I I I I I I I I I I - t r-.' t-~----IOUT rI- ton = toll = OA 61,L1 VOUT + VF (8) where: I L 1 = Filter Inductance 61, = Change in Inductor Current VOUT = Output Voltage CT = Timing capacitor Ic = Oscillator charging current 6 v = 0.5 V Typical VF = Steering Diode Forward Voltage Drop ==~---VOUT t CT6v Ic 101 VOUT (6) Whenever VOUT falls to the level specified in Equation 4, the comparator changes state and the output switches on. It remains in this state until the voltage across CT reaches a positive threshold level. The rate of CT charge is determined by the size of the timing capacitor and the magnitude of the constant current source inside the oscillator. Charging current is typically 25 /lA and discharging current is 225 /lA. From the equation describing on and off time duration, the frequency of oscillation can be deduced: I I I I ~--IOUT 1L1 (5) Equilibrium is reached at the completion of the on cycle when the comparator input has exceeded the reference level. When the comparator output goes LOW, the oscillator output is disabled and Q1 switches off. VOUT then begins to fall at a rate determined by the ratio of the output voltage to the inductor value. Since the duty cycle is dependent only upon the magnitude of the input-to-output voltage differential, it follows that variations of output voltage with load should be minimal. Basic switching regulator waveforms are shown in Figure 13. Fig~. (4) RIPPLE I ovl 5-25 Understanding the Switching Regulator Fig. 14 SH1605 Block Diagram ,----------Ql(2)------i 8 - L1 I, I I I I I lOUT 7 I + I I Rs I Co VOUT I I I I R2 Rl I .::.v 3 I IL ____________________ ~ CEXT SWITCH ON COMMAND ~ TLCOMPARATOR SWITCH OFF COMMAND ~ .tr.::.V Nominal Frequency = - - - - - - CT6v + 6 I, (nom)L 1 Ie Notes: 1. Use a 47 /-LF input capacitor to minimize switching transients. 2. Maximum T c for 5 A operation is 60°C. 3. All measurements taken relate to ground unless otherwise shown. VOUT (9) Since the required louT(min) is 1 A to maintain continuous operation, the peak-to-peak current excursion must be equal to 2 A or less, i.e., + VF 61, = 2 IOuT(min) For improved system efficiency, the operating period should always be many times longer than the device transition times. A trade off must be sought between inductor size and efficiency when selecting the frequency of operation. To calculate the value of the inductor keeping the efficiency/component-size tradeoff in mind use Equation 1. For this example ton = 60 f.LS is selected. ton is determined by the designer and depends upon the desired frequency of operation under expected constant load conditions where frequency = 1/(ton + toll). ton must always be greater than ts + td = 5.1 f.LS, typically, from the SH 1605 data sheet. Off time, tofl, is determined by the ratio of input voltage to output voltage where Design Example Figure 15 is a typical design of a step-down switching regulator using the SH1605. Nominal Design Objectives VOUT = + 5 V Line Regulation = 2% IOuT(max) = 5.0 A Load Regulation = 2% IOUT(min) = 1.0 A Ripple (max) = 0.1 Vpk-pk VIN = 12 to 18 V Efficiency = 70% ~ tofl=tonX ( -VIN --1 VOUT First, Rs is calculated from Equation 5: Rs = (2 X 103 ) (VOUT - 2.5) = 2 kf! 2.5 5-26 Understanding the Switching Regulator Fig. 15 Design Example r SHI605 + 25~F V'N 1 .OO33~F 4 50V Load Reg. ~ 50 mV (1 A :s lOUT :s 5 A) Line Reg. ~ 50 mV (12 V:s V'N:S 18 V) Circuit Performance V'N ~ 12-18 V VOUT ~ 5.06 V Note In this example the SH1605 must be mounted on a heat sink with a maximum thermal resistance of 4° C/W. (Equation 3). Thus with a known ratio of VINNoUT the designer is offered a trade-off between frequency of operation. efficiency and component size. Minimum Frequency From Equation 1: ---- = L1 = 1.7X10-4 (VIN(nOm) - VOUT) ton L':.11 10 2" (6 X ( VIN(max)L-1 VOUT) X ton Wh ere: L':. I1(max) 10- 5 ) = 5.9 kHz 300 /-LH where VIN(nOm) = 15 V, ton = 60 /-LS = 2.6A One very important element in achieving the optimum performance in a switching regulator is to insure the inductor is kept below the specified saturation limits. From Equation 1 Since the timing capacitor controls the 60 /-LS on time, CT can be determied using Equation 7: The output capacitor can now be determined as follows: CT = (ton) (Ic) = (6 X 10- 5) (2.5 X 10- 5L 3000 pF L':.v 5 X 10- 1 CO(min) = (8 f(min) Vripple(maX)) where Ie = 25 /-LA nominal per data sheet. 2 (8 X 5.9 X 103 ) X (1 X 10- 1) The final step is to determine the requirements for the output capacitor Co to obtain the desired value of ripple voltage. Consideration must be given to the absolute value of Co as well as the internal effective series resistance (ESR). Since the capacitor size is inversely proportional to the operating frequency, the lowest frequency of operation must be calculated. Minimum operating frequency can be determined by uSingL':.1 1(max) vs L':.h(nom) in Equation 9. The maximum acceptable ESR is therefore ESR(max) = Vripple(maX) L':.11(maX) 1 X 10- 1 2.6 = 0.038\1 5-27 Understanding the Switching Regulator Short Circuit Current Limit Space limitations and the already high packing density attained in the SH1605 prevent the inclusion of a short circuit current limit in the product. For those occasions where short circuit protection is required (i.e., prototype designs and lab testing), a schematic for an external protection network is shown in Figure 16. Normally, the minimum capacitance value should be increased considerably if a low ESR capacitor is not used. As a final step for minimizing switching transients at the device input, a low ESR capacitor must be used for decoupling purposes between the input terminal and ground. Conclusion The SH1605 is a highly versatile building block for high current, step-down switching regulator systems. However, to attain optimum performance and reliability the following guidelines should be followed: • Keep operating period long, relative to the device switching times, for optimum efficiency • Insure that the inductor stays out of saturation and minimize the series resistance. • Use high quality capacitors for input and output to minimize ripple and noise. Heatsink Designs While heatsinking is generally not a problem with the SH1605 due to its high efficiency, mounting of the package can have a dramatic effect on OJA. Cutting a large hole or curved slot around all eight leads leaves only the package fringes for heat transfer. A Ocs thermal resistance of 4.0 to 4.5°C/W will result from this type of mounting. A Ocs thermal resistance of 0.3 to O.4°C/W can be obtained using a hole configuration similar to Thermalloy pattern 15 or IERC pattern LAIC, UP or HP and a good thermal conducting compound. 8 Pin TO-3 Sockets Sockets are a .definite convenience when prototyping, testing and even sometimes for small volume production runs. Standard sockets are commercially available from a number of manufacturers. For a partial list of suppliers refer to the SH 1605 data sheet. Designer's Note As an aid in designing with the SH1605, 5 Amp Switching Regulator, the following is a review of several characteristics of the device which should be recognized and understood by the designer. Figure 16. Switching Regulator with Short Circuit Protection .1 ~! I-....".f\r--.......- - - - - - - - - - - VOUT 1k FDH600 6 10k 2N2222 1k 5k 5-28 Understanding the Switching Regulator grounding pOints in the system. "Input ground" is defined as the connection point between the negative side of the input filter capacitor and the incoming ground line. "Output ground" is a ground point as close to the load as possible. The input and output ground pOints are connected but distinctly separate thus minimizing system ground loops and their effect on output voltage regulation. Grounding Switching power supplies are by nature more susceptible to grounding problems than linear power supplies because of larger ripple currents. It is generally recommended that a ground plane be used. An ideal connection diagram to minimize grounding problems is shown in Figure 17. A common problem encountered with the SH1605 is excessive noise, or ripple, which is almost always generated by improper grounding. Care must be taken in the design and layout of the breadboard to eliminate any possible ground loops. This is accomplished by observing very standard layout procedures. The following diagram explicitly illustrates where the ground connections must be made to avoid potential problems. Frequency of Operation The SH1605 is a frequency modulated switcher. Thus, frequency will vary somewhat during operation depending upon power demand. When frequency is designed to fall mostly within audio ranges, users may find the continuously varying tone an annoyance. It is, therefore, recommended that users either provide for sound insulation or design for frequencies outside the normal human audio range. Pin 7, which is the anode of the steering diode and which carries up to 5 A of ripple, must be tied to "input ground" ... not the case and not "output ground". An incorrect connection here accounts for at least 80% of the field problems. To further improve system performance, the negative sides of both the timing capacitor and the decoupling capacitor should be tied together at the case with a single lead going to "output ground" and the negative side of the output filter capacitor should be connected directly to "input ground." Note that there are two distinct Although the SH 1605 is capable of operating across a broad range of frequencies, it is recommended that the user design his system to operate between 20KHz and 30KHz for optimum efficiencies and performance. For convenience, a circuit for frequency locking is shown in Figure 18. Figure 17. An Ideal Connection Diagram -+ r - RIPPLE SA RIPPLE _S 8 +C'N 1000 "F SOV 3 SH160S + 300 "H Rs' ~c ~(I\ COUT t r + V,N 2000 "F SOV VOUT + 7 COECOUPLING SA! DC RIPPLE LOAD t SINGLE POINT INPUT GROUND _SADC _SmADC _ 2S MA RIPPLE 'Metal Film Resistor or Temp. Coel. < 100ppmfOC 5-29 SINGLE POINT OUTPUT GROUND Understanding the Switching Regulator Figure 18. SH1605 Frequency Locking Network L 8 5 VOUT SH160S 3 Rs CASE 4 2 FDH600 33 k!l Notes 1. Diode FDH600 in series with 33KO resistor are the frequency locking network which facilitates measurement and minimizes noise. 2. As input to output voltage ratio is increased, the operating frequency (fo), will decrease according to the expression shown below: Vo 1 F(o) = ( - ) ( - ) where Ton Vin Ton C T 6V Ie 5-30 r Power Supply Design FAIRCHIL.D A Schlumberger Company A power supply normally operates from an ac line. This ac input voltage must be converted to unregulated dc by some form of rectifier/filter combination and then to regulated dc using a voltage regulator. This chapter discusses the performance characteristics of the most common forms of rectifier/filter combinations and provides appropriate design equations for any output voltage and current. Definition of Terms Single Phase, Half Wave Rectifier Figure 1 is a half wave rectifier and capaCitor filter. Without the capacitor, peak current is VM 1M = - - Rs + RL Parameter Definition VM peak input voltage Vo dc output voltage Vpk transformer peak voltage Vs ac input voltage F form factor of the load current: Irms/lo lac effective value of all alternating components of load current, i.e., the current reading on an ac meter on the positive half cycle (or forward conduction cycle) of the input voltage. Some additional electrical characteristics follow. Irms 10 = = 1M peak current through each rectifier average value of the load current, the reading on a dc meter 1'=1.21 Irms Po 7JR effective value of the total load current = 40.6 ~;~ ac input power % dc output power load resistance total series resistance, or the source resistance plus any added resistance plus the diode series resistance Note that for a resistive load, the maximum ripple factor is 121 % which, under most circumstances, requires filtering. When the capacitor is added across the load resistor, the ripple is reduced proportionate to the RLC product (Figure 2). ripple factor in all charts normalized as 100% equal to 1, One possible problem with any capacitive filter is the high peak current drawn due to the diode back-bias present throughout most of the input cycle. This is a result of the voltage stored across the filter capacitor. The rectifier conducts only during that short period of time when the input voltage exceeds the capacitor voltage by one diode drop. During conduction, the rectifier must supply the capaCitor with sufficient energy to hold the ripple within specification until the next conduction cycle. Figure 3 is a plot of the IM/lo ratio versus the RLC product with the Rs/RL ratio as a variable. Notice thatthe surge-todc ratio of current increases as a function of both increasing capacitor value and of a reduced sourceto-load impedance ratio. l' = I ) 2 (F2-1) = [ ( ;:s_1 ] rectification efficiency, Po X 100% Pin w 5-31 2 71'f where f = line frequency 1/2 Power Supply Design Fig. 1 Half-Wave Rectifier Circuit with Capacitive Filtering RS c rv Vs = VM sin wt Fig. 2 Ripple Factor vs WRlC 100 10 1,0 a: 0 .. ... i l- t> u.. 0.1 ~ I 910 / V l- ~ ii: 0,01 0,02 0,06 0,1 0,2 0.5 1 ;j'. I ~ 2 6 a: 10 50 100 a: ;;; 1 10 100 1000 10,000 0,1 100 1000 wRle Fig. 4 DC-to-Peak Ratio 0,05 0,5 1 2 4 6 10 1,0 0,9 0,8 0,7 12.5 rfl 15 I ~ 20 25 a: 30 C/l 40 a: 60 80 100 ::i! 0,6 ': 0,5 0 > 0,4 0,3 0,2 0,1 10 100 1000 wRLe Fig. 5 Half-Wave Rectifier RS rv Vs = VM sin wt de output-to-peak input voltage ratio approaches unity as the filter factor goes up and also as the source-to-Ioad impedance ratio decreases. Because of the relatively large value of the filter capacitor required for a given ripple factor, the use of the halfwave capacitor filter is usually limited to low current applications. When the ripple factor, load impedance, and ware known, the required capacitance can be determined from Figure 2, Because of the high turn-on surge, an external series limiting resistor is normally needed. Figure 4 is a plot of the dc-to-peak voltage ratio with the filter product as the X axis and the source/load impedance ratio as the third parameter, Note that the 5-32 Power Supply Design Half Wave Rectifier With Series Inductive Filter Figure 5 is a half-wave rectifier with series inductive filtering. The inductor, in series with the load, prevents any rapid changes in the current flow and thus reduces the ripple factor by acting as an energy storage device. When the current flow is above the average current required, energy is stored in the inductor, and when the current is below the average, the stored energy is released. Figure 6 is the plot of ripple factor versus filter product for the inductor input filter. Because of the energy storage available with an inductor, the peak current through the rectifier is little more than the average current. However, the peak inverse voltage PIV seen by the rectifier is simply VM, the peak input voltage. Figure 7 is a plot of VOIVM ratio as a function of the inductor filter product. Single-Phase Full-Wave Rectifier Figure 8, a basic full wave rectifier, has the following electrical characteristics. Irms = Po l1R 1M 21M 10 V2 1'=0.48 1r r (! VM 2RL (Rs+RL)2 81.2 ( 1+ :~) 0/0 Fig. 8 Basic Full-Wave Rectifier Fig. 6 Ripple Factor vs Filter Product 2.0 1.76 1.50 ~ 1.25 t; ~ 1.0 w to. 76 \ "'0.50 0.25 r-- o 0.1 There are two interesting features. Efficiency has doubled, as can be expected when doubling the number of rectifiers. In addition, the ripple factor has decreased from 121 % to 48% in comparison with the half-wave circuit. Even with ripple reduction, a 48% factor is normally too high to be useful and must be filtered. Figure 9 is the filter product plot for both capacitive and inductive filters, assuming Rs < < RL. High peak currrents are always associated with capacitive filters and Figure 10 plots the ratio of peak-to-dc current as a function of the filter product. The relationship between the filter product, the Rs/RL ratio and the dc output-to-peak input voltage is given in Figure 11 for the capacitive input filter. Load regulation may also be determined from Figure 11 by using the high and low limits for RL. 1.0 10 100 1000 wL/RL Fig. 7 VOIVM Ratio 0.5 0.4 :IE 0.3 > o > 0.2 1\ 0.1 o 0.1 " 10 Fig. 9 Filter Product 100 0.5 1000 ~. wL/AL 0.4 0.3 0.2 (> ~ \Cl~ 0.1 :i o I 0.1 5-33 \: \ ~ '2.~ ~ 10 100 1000 Power Supply Design Fig. 10 Peak-to-DC Ratio 1000 rl~ - :J - ii' 100 RL - c-1 i 20 - -" = Step 5 Find the transformer peak input voltage from the following. Vpk = diode forward voltage. One diode forward-voltage drop for a center-tapped fullwave input, two diode forward-voltage drops for a full-wave bridge i 1 10 0.1 100 +~ Fig. 11 Load Regulation i 1:1 ! I 0.9 ::; :: 0.6 o > 0.5 • Ie' 0.4 0.3 0.1 " using the filter values from Figure 11. 1 5 6 8 10 12.5 15 20 25 i i ~ VONM 0.05 0.1 0.5 0.8 0.7 Vpk 10 100 Vpk = 0.7 wRlC 1Q proceed with the following steps. Step 1 {= Find the filter product from Figure 9 Calculate RL 26 V peak or 52 V pk-pk or 18.6 Vrms 0.47 0.47 {= ------------------------------ (4w 2 L 1C1 -1) (4w2 L2C2 -1 )-(4w 2 Ln Cn -1) Calculate C C from Figure 11.) or, if n L-section filters are cascaded, then the ripple factor is: 20 RL = = 20[1 1 Step 3 + 25.3 = 10 4w 2 LC-1 for Step 2 = LC Section Filter The LC section filter is one method of reducing ripple levels without the need for single, large value filter omponents. The basic circuit is shown in Figure 12. As a general rule, the capacitive reactance should always be less than 10% of the load resistance at the second harmonic of the incoming frequency. All the succeeding information is based upon this ratio. The ripple factor for an L-section filter has the form: { < 0.1 = 0.82 Step 6 Check peak diode current from Figure 10. For this example at a filter product of 10, the peak current is seven times the dc current, or 7 A. 1A with Rs O.7 + -.20. - (Intersect'Ion 0 f 1000 Vo = 20V = = 5 0//0 -Rs and WRLC RL 35 40 50 60 80 100 Design Example For a full-wave circuit with the following requirements, 10 5% .;- 10 1.0 ~ Step 4 Calculate = =~ WRL 24011" 10 120 X 2011" = 1300ILF 5-34 Power Supply Design Fig. 12 LC Filter c Figure 13 is a plot of the filter factor versus the w2 LC product. The one additional requirement is continuous current flow through the inductance. This says, in effect, that there is a critical inductor size. To assure this continuous current flow, a bleeder resistor RK must be used at the filter output. The critical value of inductance is Lc = Rs Fig. 13 Filter Factor vs w2 LC eua: 0.1 :tw + Ref! . ~ 0.01 ---- 3w iO 0.001 where and 10 Bleeder current IK may be assumed to be 10% of minimum load current or, if this is not a practical value, then some reasonable minimum bleeder current is selected. Once the critical inductance is found, then the capacitor value may be determined by the following steps: Set L = 2 Lc. Determine w2 LC from Figure 13for the required ripple factor. Solve for C from w2 LC = X, where X is the product from Figure 13. 1000 10.000 For minimum power dissipation, RK should be as large as possible. In some cases, since the value of critical inductance is proportional to the value of the bleeder resistor, the selection of a high value results in an inductance too large to be practical. In this case, a swinging choke or a choke whose inductance decreases with increasing current flow is needed. The peak rectifier currents depend upon the size of the inductor selected such that if L = Lc then 1M = 2 IL and if L = 2 Lc then 1M = 1.5 IL. The transformer secondary voltage is given by Vrrns = 1.11 [ Vo 100 wile + Rs (lL(rnax) IK) ] and the minimum PIV for the rectifier is 1.57 VO(rnax) for a full-wave bridge rectifier. 5-35 Power Supply Design Design Example Full wave, single-section, choke input filter design, Vo = 50 V IK = 100 mA 10 Rs = 1A = 'Y = Step 6 Calculate voltage drop both at no load and full load Va no load = IK (Rs) = 0.1 X 10 = 1 V 1% 10 Q Va full load = (10 Step 1 Calculate RK + Rs (IO(rnax) + IK)] (50 + 10 X 1.1) Vrrns = 1.11 [Vo Vrrns = 1.11 Vrrns = 1.11 (61) Step 2 Calculate Rell RKRL(rnax) RK = 1.1 X 10 = 11 V Step 7 Calculate transformer minimum rms voltages RK = Vo = ~ = 500 Q IK 100 mA Reff = + IK) Rs + RL(rnax) Vrrns = 67.5 Vrrns 500 Q (RL(rnax) = (0) Step 8 Calculate maximum output voltage Step 3 Calculate Lc Vrrns VO(rnax) = - - IKRs 1.11 Reff + Rs _ 500 + 10 Lc = -=.c'------'-_ 3w 1130 VO(rnax)= 67.5 -0.1 X 10 = 61-1 = 60Vdc 1.11 510 :;;,,;0.5H 1130 Step 9 Calculate PIV rating required Step 4 Calculate C (See Table 4-1) PIV = (1.57) VO(rnax) 'Y = 0.01 PIV = 1.57 X 60 then, = 94 V w2LcC = 12 from Figure 13 C=~= w2LC Transformer ratios are determined from Table 1. 12 (12071')20.5 2 142 X 103 X 0.5 = 0.169 X 10- 3 = 169 ~F Step 5 Calculate 1M Since L = Lc then 1M = 2 (10 + IK) 1M = 2 X 1.1 = 2.2 A 5-36 Power Supply Design Table 1. Electrical Reference Table and Rectifier Circuit Wave Shapes Characteristic Load Average Current Through Through Rectifier IF Current Through Rectifier 1M Three Phase Star (Half-Wave) (See 1-0) 1.11 Vo 1.11 Vo 0.855 Vo 0.428 Vo 0.741 Vo 0.855 Vo 0.707 Vo 0.707 Vo 0.707 Vo 0.408 Vo 0.707 Vo 0.707 Vo Single Phase Single Phase Full Wave Center-Tap Half Wave (See 1-B) (See 1-A) RMS Resistive & 2.22 Vo Input Voltage Inductive Per Transformer Leg (V1) Capacitive 0.707 Vo Peak Inverse Voltage Per Rectifier (P & V) Single Phase Full Wave Bridge (See 1-C) Three Phase Six-Phase Double Wave Three Phase Star With Full Wave (Three Phase Interphase Diametric) Transformer Bridge (See 1-E) (See 1-F) (See 1-G) R&L 3.14 Vo 3.14 Vo 1.57 Vo 2.09 Vo 1.05 Vo 2.09 Vo 2.09 Vo C 2.00 Vo 2.00 Vo 1.00 Vo 2.00 Vo 1.00 Vo 2.00 Vo 2.00 Vo R.L. &C 1.00 10 0.5010 0.5010 0.333 10 0.333 10 0.16710 0.16710 R 3.1410 1.5710 1.5710 1.21 10 1.05 10 1.0510 0.525 10 1.0010 1.0010 1.00 10 1.0010 1.0010 0.50010 L C Depends on Size of Capacitor Transformer Total Secondary PA Sine Wave 3.49 Po 1.75 Po 1.23 Po 1.50 Po 1.05 Po 1.81 Po 1.49 Po Sq. Wave 3.14 Po 1.57 Po 1.11 Po 1.48 Po 1.05 Po 1.81 Po 1.48 Po Transformer Total Primary PA Sine Wave 3.49 Po 1.23 Po 1.23 Po 1.23 Po 1.05 Po 1.28 Po 1.06 Po Sq. Wave 1.11 Po 1.11 Po 1.21 Po 1.05 Po 1.28 Po 1.05 Po 47% 47% 17% 4% 4% 4% 1 FI 2 FI 2 FI 3 FI 6 FI 6 FI 6 FI 40.6% 81.2% 81.2% 97% 99.5% 99.5% 99.5% % Ripple Lowest Ripple Frequenc~ Conversion Efficiency 3.14 Po Sine Wave Resistive 121% Load - 5-37 Power Supply Design Table 1. Electrical Reference Table and Rectifier Circuit Wave Shapes (Cont.) SINGLE PHASE HALF WAVE (1-A) SINGLE PHASE FULL WAVE CENTER TAP (1-B) SINGLE PHASE FULL WAVE BRIDGE (1-C) THREE PHASE STAR (HALF WAVE) (1-0) Vo Vo 0----- THREE PHASE FULL WAVE BRIDGE (1-E) SIX PHASE STAR (THREE PHASE DIAMETRIC) (1-F) THREE PHASE DOUBLE WAVE WITH INTERPHASE TRANSFORMER (1-G) Vo 0------- 0-------5-38 0-------- Power Supply Design Swinging Choke LC Section Filter When designing a swinging choke section filter, the inductance required at the minimum and maximum output currents can be determined as follows. When the voltage and current levels are known, Table 2 can be used to select the optimum configuration and determine transformer and rectifier characteristics. 1. Find Lc (critical inductance) Voltage Doublers Increased dc output voltage from a transformer winding can be obtained using a voltage multiplier circuit. However, this method requires additional components, i.e., two filter capacitors, and reduces the output current. A full-wave doubler and a halfwave doubler are shown in Figure 14. The half-wave doubler is generally preferred since it has a common input and output terminal. In operation, C2 is charged on one half cycle; on the second half cycle, Cl is charged thereby summing the voltages across each capacitor. This provides a doubling effect since the output voltage is approximately twice the input voltage. + Reft Rs Lc 3w where, as before: RKRL(max) Reft RK + RL(max) 2. Find L2 (inductance at maximum load current) L2 = Rs + Reft2 3w Fig. 14. Voltage Doublers where: Reft2 HALF WAVE FULL WAVE RL(min) RK RL(min) + RK C2 Vo When Lc has been determined, then the capacitor value may be calculated as before. The condition w2Lc:S 1/4 should be avoided due to possible filter instabilities. Capacitive Input Filter Characteristics Rs/RL(min) = 0.02 WCRL(min) = 12 Table 2. Capacitive Input Filter Characteristics and Rectifier Circuit Wave Shapes Characteristic Single Phase Half Wave (See 2-A) Single Phase Full Wave Center-Tap (See 2-B) Single Phase Full Wave Bridge (See 2-C) Single Phase Full Wave Voltage Doubler (See 2-D) V1 PIV Ripple IM/Rect. IRMS/Rect. SEC VA PRIVA 0.910 Vo 2.56 Vo 0.12 Vo 7.8010 2.50 10 2.35 Po 2.35 Po 0.825 Vo 2.34 Vo .06Vo 4.7510 1.33 10 2.16 Po 3.05 Po 0.805 Vo 1.14Vo .06Vo 4.75 10 1.3310 2.16 Po 2.16 Po 0.552 Vo 1.56 Vo .09 Vo 3.0010 1.1010 1.22 Po 1.72 Po 5-39 Vo Power Supply Design Table 2. Capacitive Input Filter Characteristics and Rectifier Circuit Wave Shapes (Cont.) SINGLE PHASE FULL WAVE CENTER TAP (2-B) SINGLE PHASE HALF WAVE (2-A) c c o o SINGLE PHASE FULL WAVE VOLTAGE DOUBLER (2-D) SINGLE PHASE FULL WAVE BRIDGE (2-C) RS c c c References 1. Schade, O.H., "Analysis of Rectifier Operation," Proc IRE, July 1943, Vol. 31 #7, pp. 341-361. 4. Seeley, Samuel, "EJectron Tube Circuits," McGraw-Hili, 1958, pp. 194-230. 2. Ryder, John D., "Electronic Engineering Principles," Prentice Hall, Inc. 1947, pp. 94-125. 5. Gray, Truman, S., "Applied Electronics," John Wiley and Sons, Inc., 1954 pp. 250-277. 3. Martin, Thomas L., Jr., "Electronic Circuits," Prentice Hall, Inc., 1955, pp. 506-541. 5-40 Thermal Considerations FAIRCHILD A Schlumberger Company To realize the full capabilities of the High Current Voltage Regulator, sufficient attention must be paid to proper heat removal. For efficient thermal management, the user must rely on important parameters supplied by the manufacturer, such as junction-to-case and junction-to-ambient thermal resistance and maximum operating junction temperature. The device temperature depends on the power dissipation level, the means for removing the heat generated by this power dissipatioh and the temperature of the body (heat sink) to which this heat is removed. Thermal Evaluation of Regulators To measure thermal resistance, the difference between the junction temperature and the chosen reference temperature, case, sink or ambient, must be determined. Ambient or sink temperature measurement is straightforward. For casetemperature measurement, the device should have a sufficiently large heat sink and the power level should be close to the specified rating of the package-die combination. The case temperature can be measured by an infrared microradiometer or by using a thermocouple soldered to a point in the center of the case heat-sink interface as close to the die as practical. Figure 1 shows a simplified equivalent circuit for a typical semiconductor device in equilibrium. The power dissipation, which is analogous to current flow in electrical terms, is caused by a heat source similar to a voltage source. Temperature is analogous to voltage potential and thermal resistance to ohmic resistance. Extending the analogy of Ohm's law to Measurement of the junction temperature, unfortunately, is not as simple and involves some calibrations. There are several methods available for junction-temperature measurement; the one most commonly used is described here. Thermal Shutdown Method With this method, the thermal shutdown temperature of each device is used as the thermometer in determining the thermal resistance. The device is first heated externally, with as little internal power dissipation as practical, until it reaches thermal shutdown. Then, with the device mounted on a heat sink, the regulator is powered externally until it reaches thermal shutdown again. In some cases, the ambient of the device and its heat sink may have to be elevated sufficiently to force the regulator into shutdown. The thermal resistance of the device can then be calculated by using Thermal resistance, then, is the rise in the temperature of a package above some reference level per unit of power dissipation in that package, usually expressed in degrees in centigrade per watt. The reference temperature may be ambient or it may be the temperature of a heat sink to which the package is connected. There are several factors that affect thermal resistance including die size, the size of the heat source on the die (or substrate), dieattach material and thickness, substrate material and thickness, and package material, construction and thickness. . _ TJ-Tc ()JC--- Po where (}JC is the junction-to-case thermal resistance TJ is the measured. thermal shutdown temperature Tc is the measured case temperature Po is the power dissipated to force the device into shutdown and is equal to Fig. 1 Simplified Thermal Circuit r----------. .. POWER(P) TJ JUNCTION TEMPERATURE "JC JUNCTION-TO-CASE THERMAL RESISTANCE TC CASE TEMPERATURE (VIN - VOUT) lOUT HEAT SOURCE "cs CASE-TO-SINK THERMAL RESISTANCE T5 SINK TEMPERATURE "SA SINK-TO-AMBIENT THERMAL RESISTANCE L--_ _ _ _ _ _ _---6 + VIN 10 10 is the quiescent current of the device and can be neglected for low thermal resistance packages such as the TO-3 Heat Sink Requirements When is a heat sink necessary, and what type of a heat sink should one use? The answers to these questions depend on reliability and cost TA AMBIENT TEMPERATURE 5-41 Thermal Considerations requirements. Heat sinking is necessary to keep the operating junction temperature TJ of the regulator below the specified maximum value. Since semiconductor reliability improves as operating junction temperature is lowered, a reliability/cost compromise is usually made in the device design. lies - Case-to-heat-sink thermal resistance which for large packages, can range from about O.2°C/W to about 1°C/Wdepending on the quality of the contact between the package and the heat sink. eSA - Heat-sink-to-ambient thermal resistance, specified by heat-sink manufacturer. Thermal characteristics of voltage-regulator chips and packages determine that some form of heat sinking is mandatory whenever the power dissipation exceeds 3.2 W for the high current voltage regulator TO-3 package at 25°C ambient or lower power levels at ambients above 25°C. Maximum permissible dissipation without a heat sink is determined by P If the device dissipation Po exceeds this figure, a heat sink is necessary. The total required thermal resistance may then be calculated. To choose or design a heat sink, the designer must determine the following regulator parameters. PO(max) (VIN - Maximum power dissipation: VOUT) lOUT _ T J(max) - T A(max) O(max) IIJA + Ocs IiJA(tot) = IiJC + VIN 10 TA(max) - Maximum ambient temperature the regulator will encounter during operation. TJ(max) - Maximum operating junction temperature, specified by the manufacturer. IIJC, IIJA - Junction-to-case and junction-toambient thermal resistance values, also specified by the regulator manufacturer. (IIJA = 38°C/W max. IIJC = 2.50°C/W max). = liSA = TJ(max) - T A(max) Fig. 2 Heat Sink Material Selection Guide SURFACE AREA (BOTH SIDES OF THE HEAT SINK) SQUARE INCHES [111111111111111111111111[111111111[111111111[111111111[111111111[1111[I 11I[lliiliill[lilllllIi[1 11I[lIliliiil[ 3 4 5 6 8 10 15 20 25 30 40 50 60 80 THICKNESS 3/16" III 11111111 11111 1IIIIIIdll!IIIIIIIIII!l11I1I11I11I1I1I1 i11i 1111 I I I 1I 1I : 7 7 6 65 5 44 3 3 2 2.;·5 2 1 3/32,,1111111111111111111[111111111[111 11 111111111111111 11111 11111 1 3/16" II III Illilliilllllllililillililililllllillilil i1iill II II III 1 7 THICKNESS 6 5 4 3 2 1.5 6 5 4 . 3 2.5 IlIlilllllllllllllllllllllllllllllllllllllllllll 1 7 3/32" I I 2 I 3/16" 1"I'11T'T1I"11"1I"I TTII I1Im'l irml lnTlIl1T1'l lm'1I1"'l m lil"rI"III"1'I"rI"'1"'1"r1"'1'I'II""II""II""II'T'MII1 COPPER; HORIZONTALLY· MOUNTED COPPER. VERTICALLY· MOUNTED n1 7 THICKNESS 3/32" THICKNESS PD Case-to-sink and sink-to-ambient thermal resistance information on commercially available heat sinks is normally provided by the heat sink manufacturer. A summary of some commercially available heat sinks is shown in Table 1. However, if a chassis or other conventional surface is used as a heat sink, Figure 2 can be used as a guide to estimate the required surface area. 6 5 4 3 2.5 2 8 7 6 5 4 3.5 3 2.5 2 111111111I11I1Il!11I11I1I11111I11I11111111 I I I I I 1I11 111I111Ii! 3/16" 11"11111111111111111"111""111111111111 I I I I 11111 7 6 5 4 3 2.5 2 7 6 5 4 3.5 3 2.5 2 3/32,,111111111111111111111111111111111111111111111111111 THERMAL RESISTANCE IN To determine either area required or thermal resistance of a given area, draw a vertical line between the top (or area) line down to the material of interest. 5-42 °c/w ALUMINUM. HORIZONTALLY· MOUNTED ALUMINUM. VERTICALLY· MOUNTED Thermal Considerations Table 1 Heat Sink Selection Guide This list is only representative. No attempt has been made to provide a complete list of all heat sink manufacturers. All values are typical as given by manufacturer or as determined from characteristic curves supplied by manufacturer. How to Choose a Heat Sink - Example Determine the heat sink required for a regulator which has the following system requirements: Operating Maximum Maximum Maximum ambient temperature range: 0°C-40°C junction temperature: 125°C output current: 3 A input to output differential: 5 V (JSA (JJC = 2.5°C/W maximum (from data sheet) (JJA(tot) = (JCS (JJC + (JSA = Assuming PD (Jcs = .16°C/W = then 3.16°C/W (JSA = Manufacturer and Type 0.4 (9" length) Thermalloy (Extruded) 6590 Series Thermalloy (Extruded) 6660, 0.4 - 0.5 (6" length) 6560 Series Wakefield 400 Series 0.56 - 3.0 0.6 (7.5" length) Thermalloy (Extruded) 6470 Series Thermalloy (Extruded) 6423, 6443, 0.7 - 1.2 (5 - 5.5" length) 6441, 6450 Series Thermalloy (Extruded) 6427, 6500, 1.0 - 5.4 6123, 6401, 6403, 6421, 6463, (3" length) 6176,6129,6141,6169,6135, 6442 Series 1.9 IERC E2 Series (Extruded) 2.1 IERC E1, E3 Series (Extruded) 2.3 - 4.7 Wakefield 600 Series 4.2 IERC HP3 Series 4.5 Staver V3-5-2 Thermalloy 6001 Sries 4.8 - 7.5 5-6 IERG HP3 Series Thermalloy 6013 Series 5 - 10 Staver V3-3-2 5.6 Wakefield 680 Series 5.9 - 10 Wakefield 390 Series 6 6.4 Staver V3-7-224 6.5 - 7.5 IERG Up SEries 8 Staver V1-5 8.1 Staver V3-5 8.8 Staver V3-7-96 9.5 Staver V3-3 9.5 - 10.5 IERG LA Series 9.8 - 13.9 Wakefield 630 Series 10 Staver V1-3 11 Thermalloy 6103, 6117 Series TJ - TA + (Jcs + (JSA = -125 - 40 - 2.5 3X5 Approx. (OC/W) For this example assume the IlA78HGA, 5 Amp Positive Adjustable High Current Voltage Regulator has been selected. 3°C/W This thermal resistance value can be achieved by using either 22 square inches of 3/16 inch thick vertically mounted aluminum (Figure 2) or a commercial heat sink (Table 1). Tips for Better Regulator Heat Sinking Avoid placing heat-dissipating components such as power resistors next to regulators. Keep lead lengths to a minimum and use the largest possible area of the printed board traces or mounting hardware to provide a heat dissipation path for the regulator. Be sure the heat sink surface is flat and free from .ridges or high spots. Check the regulator package for burrs or peened-over corners. Regardless of the smoothness and flatness of the package and heatsink contact, air pockets between them are unavoidable unless a lubricant is used. Therefore, for good thermal conduction, use a thin layer of thermal lubricant such as Dow Corning DC-340, General Electric 662 or Thermacote by Thermalloy. If the regulator is mounted on a heat sink with fins, the most efficient heat transfer takes place when the fin is in a vertical plane, as this type of mounting forces the heat transfer from fin to air in a combination of radiation and convection. If it is necessary to bend any of the regulator leads, handle them carefully to avoid straining the package. Furthermore, lead bending should be restricted since repeated bending will fatigue and eventually break the leads. 5-43 FAIRCHILD A Schlumberger Company Fairchild Field Sales Offices, Representatives and Distributors 6-2 FAIRCHILD A Schlumberger Company Alabama Hall Mark Electronics 4900 Bradford Drive Huntsville, Alabama 35807 Tel: 205-837-8700 TWX: 810-726-2187 Hamilton/Avnet Electronics 4692 Commercial Drive Huntsville, Alabama 35805 Tel: 205-837-7210 TWX: 810-726-2162 Arizona Hamilton/Avnet Electronics 505 South Madison Drive Tempe, Arizona 85281 Tel: 602-231-5100 TWX: 910-950-0077 Kierulff Electronics 4134 East Wood Street Phoenix, Arizona 85040 Tel: 602-243-4101 Wyle Distribution Group 8155 North 24th Avenue Phoenix, Arizona 85021 Tel: 602-249-2232 TWX: 910-951-4282 California Anthem Electronics, Inc. 21730 Nordhoff Street Chatsworth, California 91311 Tel: 213-700-1000 TWX: 910-493-2083 Anthem Electronics, Inc. 4125 Sorrento Valley Blvd. San Diego, California 92121 Tel: 714-279-5200 Anthem ElectroniCs, Inc. 174 Component Drive San Jose, California 95131 Tel: 408-946-8000 Anthem Electronics, Inc. 2661 Dow Avenue Tustin, California 92680 Tel: 714-730-8000 Arrow Electronics 9511 Ridge Haven Court San Diego, California 92123 Tel: 714-565-4800 TWX: 910-335-1195 Arrow Electronics 521 Weddell Avenue Sunnyvale, California 94086 Tel: 408-745-6600 TWX: 910-339-9371 Avnet Electronics 340 McCormick Avenue Costa Mesa, California 92626 Tel: 714-754-6111 (Orange County) 213-558-2345 (Los Angeles) TWX: 910-595-1928 Bell Industries Electronic Distributor Division 1161 N. 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Albuquerque, New Mexico 87106 Tel: 505·243·4566 TWX: 910·889·1679 Franchised Distributors United States and Canada Summit Distributors, Inc. 916 Main Street Buffalo, New York 14202 Tel: 716·884·3450 TWX: 710·522·1692 Pennsylvania Arrow Electronics 650 Seco Road Monroeville, Pennsylvania 15146 Tel: 412·856·7000 Bell Industries 11728 Linn Avenue N.E. Albuquerque, New Mexico 87123 Tel: 505·292·2700 TWX: 910·989·0625 North carolina Arrow Electronics 938 Burke Street Winston·Salem, North Carolina 27102 Tel: 919·725·8711 TWX: 510·931-3169 Pioneer Electronics 261 Gibraltar Road Horsham, Pennsylvania 19044 Tel: 215·674·4000 TWX: 510·665·6778 Hamilton/Avnet Electronics 2524 Baylor Drive, S.E. Albuquerque, New Mexico 871 06 Tel: 505·765·1500 TWX: 910·989·0614 Hall Mark Electronics 1208 Front Street, Bldg. K Raleigh, North Carolina 27609 Tel: 919·823·4465 TWX: 510·928·1831 Pioneer Electronics 259 Kappa Drive Pittsburgh, Pennsylvania 15238 Tel: 412·782·2300 TWX: 710·795·3122 New York Arrow Electronics 900 Broadhollow Road Farmingdale, New York 11735 Tel: 516·694·6800 TWX: 510·224·6155 & 510·224·6126 Hamilton/Avnet Electronics 2803 Industrial Drive Raleigh, North Carolina 27609 Tel: 919·829·8030 TWX: 510·928·1836 Schweber Electronics 101 Rock Road Horsham, Pennsylvania 19044 Tel: 215·441·0600 TWX: 510·665·6540 Pioneer Electronics 103 Industrial Drive Greensboro, North Carolina 27406 Tel: 919·273·4441 Texas Arrow Electronics 13715 Gamma Road Dallas, Texas 75234 Tel: 214·386·7500 TWX: 910·860·5377 Arrow Electronics 20 Oser Avenue Hauppauge, New York 11787 Tel: 516·231-1000 Arrow Electronics P.O. Box 370 7705 Maltlage Drive Liverpool, New York 13088 Tel: 315·652·1000 TWX: 710·545·0230 Components Plus, Inc. 40 Oser Avenue Hauppauge, New York 11787 Tel: 516·231·9200 TWX: 510·227·9869 Hamilton/Avnet Electronics '5 Hub Drive Melville, New York 11746 Tel: 516·454·6000 TWX: 510·224·6166 Hamilton/Avnet Electronics 333 Metro Park Rochester, New York 14623 Tel: 716-475·9130 TWX: 510·253·5470 Hamilton/Avnet Electronics 16 Corporate Circle E. Syracuse, New York 13057 Tel: 315·437·2642 TWX: 710·541-1560 Harvey Electronics (mailing address) P.O. Box 1208 Binghamton, New York 13902 (shipping address) 1911 Vestal Parkway East Vestal, New York 13850 Tel: 607·748·8211 Harvey Electronics 60 Crossways Park West Woodbury, New York 11797 Tel: 516·921·8920 TWX: 510·221·2184 Schweber Electronics Jericho Turnpike Westbury, LI., New York 11590 Tel: 516·334·7474 TWX: 910·222·3660 Ohio Arrow Electronics 7620 McEwen Road Centerville, Ohio 45459 Tel: 513·435·5563 TWX: 810·459·1611 Arrow Electronics 10700 Corporate Drive, Suite 100 Stafford, Texas 77477 Tel: 713·491·4100 TWX: 910·880·4439 Arrow Eiectronics 6238 Cochran Road Solon, Ohio 44139 Tel: 216·248·3990 TWX: 810·427·9409 Hall Mark Electronics 12211 Technology Blvd. Austin, Texas 78759 Tel: 512·258·8848 Hamilton/Avnet Electronics 954 Senate Drive Dayton, Ohio 45459 Tel: 513-433·0610 TWX: 810·450·2531 Hall Mark Electronics 11333 Page Mill Drive Dallas, Texas 75243 Tel: 214·343·5000 TWX: 910·867·4721 Hami~on/Avnet Electronics 4588 Emery Industrial Parkway Warrensville Heights, Ohio 44128 Tel: 216·831·3500 TWX: 810·427·9452 Hall Mark Electronics 8000 Westglen Houston, Texas 77063 Tel: 713·781·6100 Pioneer Electronics 4800 E. 131st Street Cleveland, Ohio 44105 Tel: 216·587·3600 Hamilton/Avnet Electronics 2401 Rutland Drive Austin, Texas 78758 Tel: 512·837·8911 TWX: 910·874·1319 Pioneer Electronics 4433 Interpoint Blvd. Dayton, Ohio 45424 Tel: 513·236·9900 TWX: 810·459·1622 Hamilton/Avnet Electronics 8750 Westpark Houston, Texas 77063 Tel: 713·780·1771 TWX: 910·881·5523 Schweber Electronics 23880 Commerce Park Road Beachwood, Ohio 44122 Tel: 216-464·2970 TWX: 810·427·9441 Hamilton/Avnet Electronics 2111 W. Walnut Hill Lane Irving, Texas 75062 Tel: 214·659·4111 TWX: 910·860·5929 Oklahoma Hall Mark Electronics 5460 S. 103rd East Avenue Tulsa, Oklahoma 74145 Tel: 918·665·3200 TWX: 910·845·2290 Schweber Electronics, Inc. 4202 Beltway Drive Dallas, Texas 75234 Tel: 214·661·5010 TWX: 910·860·5493 Oregon Hamilton/Avnet Electronics 6024 S.W. Jean Road Building C, Suite 10 Lake Oswego, Oregon 97034 Tel: 503·635·8157 TWX: 910·455·8179 6·5 Schweber Electronics, Inc. 10625 Richmond, Suite 100 Houston, Texas 77042 Tel: 713·784·3600 TWX: 910·881·4836 FAIRCHILD A Schlumberger Company Sterling Electronics 4201 Southwest Freeway Houston, Texas 77027 Tel: 713-627-9800 TWX: 910-881-5042 Telex: STELECO HOUA 77-5299 Utah Bell Industries 3639 West 2150 South Salt Lake City, Utah 84120 Tel: 801-972-6969 TWX: 910-925-5686 Franchised Distributors United States and Canada Wisconsin Hamilton/Avnet Canada Ltd. 2670 Sabourin Street SI. Laurent, Quebec, H4S 1M2, Canada Tel: 514-331-6443 TWX: 610-421-3731 Hall Mark Electronics 9657 South 20th Street Oakcreek, Wisconsin 53154 Tel: 414-761-3000 Hamilton/Avnet Electronics 2975 South Moorland Road New Berlin, Wisconsin 53151 Tel: 414-784-4510 TWX: 910-262-1182 Hamilton/Avnet Electronics 1585 West 2100 South Salt Lake City, Utah 04119 Tel: 801-972-2800 TWX: 910-925-4018 Canada Future Electronics Inc. 4800 Dufferin Street Downsview, Ontario M3H 5S8, Canada Tel: 416-663-5563 Washington Arrow Electronics 14230 N.E. 21st Street Bellevue, Washington 98005 Tel: 206-643-4800 TWX: 910-443-3033 Future Electronics Inc. Baxter Center 1050 Baxter Road Ottawa, Ontario, K2C 3P2, Canada Tel: 613-820-8313 Hamilton/Avnet Electronics 14212 N.E. 21st Street Bellevue, Washington 98005 Tel: 206-453-5844 TWX: 910-443-2469 Future Electronics Inc. 237 Hymus Blvd. Pointe Clare (Montreal), Quebec, H9R 5C7, Canada Tel: 514-694-7710 TWX: 610-421-3251 Radar Electronic Co., Inc. 168 Western Avenue W. Seattle, Washington 98119 Tel: 206-282-2511 TWX: 910-444-2052 Hamilton/Avnet Canada Ltd. 6845 Rexwood Road, Units 3-4-5 Mississauga, Ontario, L4V 1R2, Canada Tel: 416-677-7432 TWX: 610-492-8867 Wyle Distribution Group 1750 132nd Avenue N.E. Bellevue, Washington 98005 Tel: 206-453-8300 TWX: 910-444-1379 Hamilton/Avnet Canada Ltd. 210 Colonnade Road Nepean, Ontario K2E 7L5, Canada Tel: 613-226-1700 Tlx: 0534-971 6-6 Semad Electronics Ltd. 620 Meloche Avenue Dorval, Quebec, H9P 2P4, Canada Tel: 604-299-8866 TWX: 610-422-3048 Semad Electronics Ltd. 105 Brisbane Avenue Downsview, Ontario, M3J 2K6, Canada Tel: 416-663-5670 TWX: 610-492-2510 Semad Electronics Ltd. 864 Lady Ellen Place Ottawa, Ontario Kl Z 5M2, Canada Tel: 613-722-6571 TWX: 610·562-1923 A Schlumberger Company Sales Representatives United States and Canada California Nevada Utah Magna Sales, Inc, 3333 Bowers Avenue, Suite 295 Santa Clara, California 95051 Tel: 408-727-8753 TWX: 910-338-0241 Magna Sales, Inc, 4560 Wagon Wheel Road Carson City, Nevada 89701 Tel: 702-883-1471 Simpson Associates, Inc. 7324 South 1300 East, Suite 350 M idyale, Utah 84047 Tel: 801-566-3691 TWX: 910-925-4031 New York WaShington Tri-Tech Electronics, Inc. 3215 E. Main Street Endwell, New York 13760 Tel: 607-754-1094 TWX: 510-252-0891 Magna Sales, Inc. FAIRCHIL.D Colorado Simpson Associates, Inc. 2552 Ridge Road Littleton, Colorado 80120 Tel: 303-794-8381 TWX: 910-935-0719 Illinois Micro Sales, Inc, 54 W, Seegers Road Arlington Heights, Illinois 60005 Tel: 312-956-1000 TWX: 910-222-1833 Maryland Delta III Associates 1000 Century Plaza, Suite 224 Columbia, Maryland 21044 Tel: 301-730-4700 TWX: 710-826-9654 Massachusetts Spectrum Associates, Inc. 109 Highland Avenue Needham, Massachusetts 02192 Tel: 617-444-8600 TWX: 710-325-6665 Missouri Micro Sales, Inc, 514 Earth City Plaza, Suite 314 Earth City, Missouri 63045 Tel: 314-739-7446 Tri-Tech Electronics, Inc. 590 Perinton Hills Office Park Fairport, New York 14450 Tel: 716-223-5720 TWX: 510-253-6356 Tri-Tech Electronics, Inc. 6836 E. Genesee Street Fayetteville, New York 13066 Tel: 315-446-2881 TWX: 710-541-0604 Tri-Tech Electronics, Inc, 19 Davis Avenue Poughkeepsie, New York 12603 Tel: 914-473-3880 TWX: 510-253-6356 Oregon Magna Sales, Inc. 8285 S.W. Nimbus Avenue, Suite 138 Beaverton, Oregon 97005 Tel: 503-641-7045 TWX: 910-467-8742 6-7 Benaroya Business Park Building 3, Suite 115 300 120th Avenue, N.E, Bellevue, Washington 98004 Tel: 206-455-3190 Wisconsin Larsen Associates 10855 West Potter Road Wauwatosa, Wisconsin 53226 Tel: 414-258-0529 TWX: 910-262-3160 A Schlumberger Company Sales Offices United States and Canada Alabama Huntsville Office 500 Wynn Orive, Suite 511 Huntsville, Alabama 35805 Tel: 205-837-8960 Indiana Ft. Wayne Office 2118 Inwood Drive, Suite 111 Ft. Wayne, Indiana 46815 Tel: 219-483-6453 TWX: 810-332-1507 North Carolina Raleigh Office 1100 Navaho Drive, Suite 112 Raleigh, North Carolina 27609 Tel: 919-876-9643 Arizona Indianapolis Office 7202 N, Shadeland, Room 205 Castle Point Indianapolis, Indiana 46250 Tel: 317-849-5412 TWX: 810-260-1793 Ohio Dayton Office 5045 North Main Street, Suite 105 Day1on, Ohio 45414 Tel: 513-278-8278 TWX: 810-459-1803 Kansas Kansas City Office 8600 West 11 Oth Street, Suite 209 Overland Park, Kansas 66210 Tel: 913-649-3974 Oklahoma Tulsa Office 9810 East 42nd Street, Suite 127 Tulsa, Oklahoma 74145 Tel: 918-627-1591 Maryland Columbia Office 1000 Century Plaza, Suite 225 Columbia, Maryland 21044 Tel: 301-730-1510 TWX: 710-826-9654 Oregon Portland Office 8285 S.W. Nimbus Avenue, Suite 138 Beaverton, Oregon 97005 Tel: 503-641-7871 TWX: 910-467-7842 Massachusetts Framingham Office 5 Speen Street Framingham, Massachusetts 01701 Tel: 617-872-4900 TWX: 710-380-0599 Pennsylvania Philadelphia Office' 2500 Office Center 2500 Maryland Road Willow Grove, Pennsylvania 19090 Tel: 215-657-2711 FAIRCHIL.C Phoenix Office 2255 West Northern Road, Suite B112 Phoenix, Arizona 85021 Tel: 602-864-1000 TWX: 910-951-1544 California Los Angeles Office' Crocker Bank Bldg, 15760 Ventura Blvd" Suite 1027 Encino, California 91436 Tel: 213-990-9800 TWX: 910-495-1776 San Diego Office' 4355 Ruffin Road, Suite 100 San Diego, California 92123 Tel: 714-560-1332 Santa Ana Office' 1570 BroOkholiow Drive, 'Suite 206 Santa Ana, California 92705 Tel: 714-557-7350 TWX: 910-595-1109 Santa Clara Office' 3333 Bowers Avenue, Suite 299 Santa Clara, California 95051 Tel: 408-987-9530 TWX: 910-338-0241 Colorado Denver Office 7200 East Hampden Avenue, Suite 206 Denver, Colorado 80224 Tel: 303-758-7924 Connecticut Danbury Office 57 North Street, #206 Danbury, Connecticut 06810 Tel: 203-744-4010 Florida Ft. Lauderdale Office Executive Plaza, Suite 112 1001 Northwest 62nd Street Ft. Lauderdale, Florida 33309 Tel: 305-771-0320 TWX: 510-955-4098 Orlando Office' Crane's Roost Office Park 399 Whooping Loop Altamonte Springs, Florida 32701 Tel: 305-834-7000 TWX: 810-850-0152 Georgia Atlanta Sales Office Interchange Park, Bldg, 2 4183 N,E, Expressway Atlanta, Georgia 30340 Tel: 404-939-7683 Illinois Itasca Office 500 Park Blvd" Suite 575 Itasca, Illinois 60143 Tel: 312-773-3300 Michigan Detroit Office' 21999 Farmington Road Farmington Hills, Michigan 48024 Tel: 313-478-7400 TWX: 810-242-2973 Minnesota Minneapolis Office' 4570 West 77th Street, Room 356 Minneapolis, Minnesota 55435 Tel: 612-835-3322 TWX: 910-576-2944 New Jersey New Jersey Office Vreeland Plaza 41 Vreeland Avenue Totowa, New Jersey 07511 Tel: 201-256-9006 New Mexico Albuquerque Office North BUilding 2900 Louisiana N,E. South G2 Albuquerque, New Mexico 87110 Tel: 505-884-5601 TWX: 910-379-6435 New York Fairport Office 815 Ayrault Road Fairport, New York 14450 Tel: 716-223-7700 Melville Office 275 Broadhollow Road, Suite 219 Melville, New York 11747 Tel: 516-293-2900 TWX: 510-224-6480 Poughkeepsie Office 19 Davis Avenue Poughkeepsie, New York 12603 Tel: 914-473-5730 TWX: 510-248-0030 , Field Application Engineer 6-8 Tennessee Knoxville Office Executive Square 1/ 9051 Executive Park Drive, Suite 502 Knoxville, Tennessee 37923 Tel: 615-691-4011 Texas Austin Office 8240 Mopac Expressway, Suite 270 Austin, Texas 78759 Tel: 512-837-8931 Dal/as Office 1702 North Col/ins Street, Suite 101 Richardson, Texas 75081 Tel: 214-234-3391 TWX: 910-867-4757 Houston Office 9896 Bissonnet-2, Suite 470 Houston, Texas 77036 Tel: 713-771-3547 TWX: 910-881-8278 Canada Toronto Regional Office 2375 Steeles Avenue West, Suite 203 Downsview, Ontario M3J 3A8, Canada Tel: 416-665-5903 TWX: 610-491-1283 Franchised Distributors I=AIRCHILD A Schlumberger Company Austria BVG elektrot. bauelemente vertriebsges.mbH Rottstr. 8-1 0 1140Wien Tel: (0043) 0222/949373 TWX: 135123 Brazil Alfatronic Imp Exp Repres Llda. Av. Repousas, 1498 - Sao Paulo, Brazil Tel: (011) 852-8277 Comercial Radio Car Llda. Av. Alberto Bins, 615 - Porto Alegre, Brazil Tel: (0512) 25-8879 Datatronix Eletr Ltda. Av. Pacaembu, 746 - Sao Paulo, Brazil Tel: (011) 826-0111 Eletropan Imp Repres Llda. R. Barra Bonita, 18 - Sao Paulo, Brazil Tel: (011) 295-0293 Intertek Comp Eletr Llda. R. Tagipuru, 235 - 80A Tel: (011) 67-0582 Karimex Imp Exp Llda. Rua Guararapes, 1826 Tel: (011) 241-2814 Sao Paulo, Brazil Sao Paulo, Brazil Semicon Sem E Comp Eletr Llda. R. Coronel Oscar Porto, 841 - Sao Paulo, Brazil Denmark E Friis Mikkelsen AS 51 Krogshojvej DK2880 Bagsvaerd, Denmark Tel: (02) 986333 TWX: 37350 Multikomponent (Standard Electric) AS Fabriksparken 31 DK 2600 Glostrup, Denmark Tel: (02) 456645 TWX: 33355 Finland Multikomponent Kuortaneenkatu 1 SF-00520 Helsinki 52, Finland Tel: 009358/073 91 00 TWX: 12 1450 France Almex 48, Rue De L' Aubepine B.P.102 Tel: 666-21-12 TWX: 250.067 Aufray Centre De Gros Zone Industrielle 76800 St Etienne Du Rouvray Tel: (35) 65-22-22 TWX: 180.503 Bellion Electronique Z.I. Kerscao Brest B.P.16 29219 Le Relecq Kerhuon Tel: (98) 28-03-03 TWX: 940.930 Dimex (Stockiste) 12, Rue Du Seminaire 94516 Rungis Cedex Tel: 686.52.10 TWX: 200.420 Feutrier Avenue Trois Clorieuses 42270 St Priest En Jarez Tel: (77) 74-67-33 Feutrier lie De France 8, Rue Benoit Malon 92150 Suresnes Tel: 772-46-46 TWX: 610.237 Gros Electronique 13, Avenue Victor Hugo B.P.63 59350 St Andre Lez Lille Tel: (20) 51.21.33 TWX: 120.257 Paris Sud 1 Route De Champlan 91300 Massy Tel: 920-66-99 R.E.A. 9, Rue Ernest Cognacq B.P.5 92300 Levallois Tel: 758-11-11 TWX: 620.630 S.C.T. (Toutelectric) 15, Boulevard Bon Repos B.P.406 31008 Toulouse Tel: (61) 62.11.33 TWX: 531.501 Scientech 11, Avenue Ferdinand Buisson 75016 Paris Tel: 609-91-36 TWX: 260.042 S.R.D. Chemin Des Pennes Au Pin Plan De Campagne 13170 Les Pennes Mirabeau Tel: (42) 02.91.08 TWX: 440.076 Germany Astek GmbH Carl-Zeiss-Str.3 2085 Quickborn Tel: (0049) 04106171 084 TWX: 0214082 Dr. G. Dohrenberg Bayreuther Str. 3 1000 Berlin 30 Tel: (0049) 030/2138043 TWX: 0184860 E2000 Vertriebs GmbH Neumarkter Str. 75 8000 Manchen 80 Tel: (0049) 089/434061 TWX: 0522561 ElcowaGmbH Str. der Republik 17-19 6200 Wiesbaden Tel: (0049) 06121/65005 TWX: 04186202 IBH Gutenbergring 35 2000 Norderstedt Tel: (0049) 040/5231933 TWX: 02174188 ProtecGmbH Franz-Liszt-Str. 4 Tel: (0049) 089/603006 TWX: 0529298 6-9 International Positron Bauelem. Vertriebs GmbH Benzstr.1 7016 Gerlingen Tel: (0049) 07156/23051 TWX: 07245266 Spezial Electronic KG Kreuzbreite 15 3062 Backeburg Tel: (0049) 05722/2030 TWX: 0971624 T echnoprojekt Heinrich-Baumann-Str.30 7000 Stuttgart Tel: (0049) 0711/280281 TWX: 0721437 Italy Region Of Campania: A.E.P. Via Terracina. 311 - 80125 Napoli Tel: 081-630006 TWX: 721129 Region Of Emilia Romagna: Adelsy S.A.S. Via Lombardia. 17/2 - 40139 Bologna Tel: 051-540150 TWX: 510226 Adelsy Hellis P. ZZA Amendola. 1 - 41049 Sassuolo (MO) Tel: 059/8041 04-864990 Region Of Lazio: Pantronic S.R.L. Via Flaminia Nuova. 219 00191 Roma Tel: 06/3284866-3288048 TWX: 612405 Pantron Region Of Lombardia: Claitron S.P.A. Viale Certosa 269 - 20151 Milano Tel: 3088063/5/7/-3087330-3088506 3088030-306539-305580 TWX: 313843 Claimi Comprel S.R.L. Viale Romagna. 1 20092 Cinisello Balsamo (MI) Tel: 6120641 TWX: 332484 Kontron S.P.A. Via Fantoli. 16/15 - 20138 Milano Tel: 50721 TWX: 315430 Kontmi I Region Of Marche: Comprel S.R.L. Traversa Carlo Moderno. 24 Casella Postale 9 60025 Loreto (AN) Tel: (071) 977693 Region Of Piemonte: Claitron S.P.A. Via Tazzoli. 158 10137 Torino Tel: 011/3097173/306540 Pantronic S.R.L. Via Crevacuore. 65 10146 Torino Tel: 011-790079-795981 TWX: 221420 A Schlumberger Company Franchised Distributors Region Of 3 Venezie: Comprel S.R.L. Via V. Veneto. 33 36100 Vicenza Tet: 0444-26912 Hakou Corp. Daishin Bldg. Gokisho-Dohri. Showa-Ku Nagoya-Shi. Aichi 466. Japan Tel: (052) 853-5621 Kontron S.P.A. Via Forcellini. 4 35100 Padova Tel: 049/754717/850377 Hamilton Avnet Electronics Corp. Nishi-Honmachi Zennikku Bldg. 10-10. Nishi-Honmachi 1-Chome. Nishi-Ku Osaka 550. Japan Tel: (06) 533-5855 FAIRCHIL.D Japan Alpha Denshi Corp. Yamajin Bldg. 1-11. Esaka-Cho 2-Chome. Suita-Shi Osaka 564. Japan Tel: (06) 384-2281 Asahi Glass Corp. Hankyu Terminal Bldg. 1-4. Shibata 1-Chome. Kita-Ku Osaka 530. Japan Tel: (06) 373-5895 Asahl Glass Corp. Kishimoto Bldg. 2-1. Marunouchi 2-Chome. Chiyoda-Ku Tokyo 100. Japan Tel: (03) 218-5800 Ashitate Denki Corp. Higashi-Nagasaki Bldg. 1-14. Kanda-Iwamoto-Cho. Chiyoda-Ku Tokyo 1D1 .. Japan Tel: (03)255-5151 Ashltate Denki Corp. Highness-Katahira 901 3-36. Katahira 1-Chome. Sendai-Shi Miyagi 980. Japan T91:(0222) 66-8951 Dairiichi Seigyo Kiki Corp. Kouraku BI~g. 1-8. Kouraku 1-Chome. Bunkyo-Ku TOkyo 112. Japan Tel: (03) 811,-9205 Fuji Electronics Corp. Fusou Bldg. Hamilton Avnet Electronics Corp. Yu and You Bldg. 1-4. Nihonbashi Horidome-Cho. Chuo-Ku Tokyo 103. Japan Tel: (03) 662-9911 lnaba Sangyo Kiki Corp. 4-6. Honda 1-Chome. Nishi-Ku Osaka 550. Japan Tel: (06) 582-8483 Kanematsu Semiconductor Corp. Bingo-Cho Nomura Bldg. 2-5. Bingo-Cho. Higashi-Ku Osaka 541. Japan Tel: (06) 222-1851 Kanematsu Semiconductor Corp. Daini-Nagaoka Bldg. 8-5. Hatchobori 2-Chome. Chuo-Ku Tokyo 104. Japan Tel: (03) 552-6091 Nakamura Denki Corp. 3-5. Soto-Kanda 1-Chome. Chiyoda-Ku Tokyo 101. Japan Tel: (03) 255-6831 Okamoto Musen Denki Corp. 7-28. Hatae Dohri. Nakarnura-Ku Nagoya-Shi. Aichi 453. Japan Tel: (052) 461-4111 Okamoto Musen Denki Corp. 2-7. Ohsumi 1-Chome. Higashi-Yodogawa-Ku Osaka 533. Japan Tel: (06) 327-1133 , 5-3. Nishi,tlonmallhi 1-Chome. Nishi-Ku Osaka 550. Japan Tel: (06)541-7112 Fuji Electronics Corp. New-Kourakuen Bldg. 22-3. Hongo 1-Chome. Bunkyo-Ku Tokyo 113. Japan Tel: (03) 815-0830 Futaba Danki Corp. SudOu Bldg. 3-9. Shimizu l-Chome Matsumoto~Shi. Nagano 390 Japan Tet: (02~)35-2329 i"utaba D8nki Corp. ShUWIl-1BR Bldg. 5-7. KOhjimachi. Chiyoda-Ku Tokyo 102. Japan Tel: (03) 230-2171 Okamoto Musen Denki Corp. 2-17. Nozawa 3-Chome. Setagaya-Ku Tokyo 154. Japan Tel: (03) 412-8211 International Radio Industrial Del Norte. S.A. Calle Del Cerro No. 18 H. Del Parral. Chih. Tel: 2-18-38 The Netherlands Inelco Components and Systems bv Turfstekerstraat 63 1431 GD-Aalsmeer Tel: (0031) 02977/28855 TWX: 14693 Rodelco Electronics Verrijn Stuartlaan 29 2280 AG-RijSwijk ZH Tel: (0031) 070/995750 TWX: 32506 Rodelco Electronics Rue de Genave 4 1140 Bruxelles Tel: (0032) 02/2166330 TWX: 61415 Norway Datamatik NS Jerikoveien 16 Oslo 10 Norway Tel: 00947/230 17 30 TWX: 16967 Sweden In Multikomponent Box 1330 S-17125Solna Sweden Tel: 00946/8830020 TWX: 10516 Norqvist + Berg Box 9145 S-102 72 Stockholm Sweden Tel: 00946/869 04 00 TWX: 10407 Switzerland MoorAG Bahnstr.58 8105 Regensdorf/Zurich Tel: (0041) 01/8406644 TWX: 0045-52042 PrimotecAG Wettinger Str. 23 5400 Baden Tel: (0041) 056/265262 TWX: 0045-58949 Osaka T okiwa Shoko 13-3. Nipponbashi 5-Chome. Naniwa-Ku Osaka 556. Japan Tel: (06) 643-3521 United Kingdom Barlec Ltd. Foundry Lane Horsham Sussex RH 13 5PX Tei: Horsham (0403) 51881 TWX: 877222 Mexico Dicopel S.A. Augusto Rodin No. 20 Mexico 18 D.F. Tel: 687-18-00 Celdis Ltd. 37/39 Loverock Road Reading Berkshire RG3 1DZ Tel: Reading (0734) 585171 TWX: 848370 Distele S.A. Obrero Mundila 736 Mexico 13 D.F. Tel: 538-05-00 Comway Electronics Ltd. Market Street Bracknell Berkshire RG12 1QP Tel: Bracknell (0344) 24765 TWX: 847201 Proveedora Electronica S.A. Prolongacion Moctezuma Ote. No. 24 Mexico 21 D.F. Tel: 554-83-00 6-10 In Electronic Services Edinburgh Way Harlow CM20 2DF Tel: Harlow (0279) 26777 TWX: 81525 FAIRCHILD A Schlumberger Company Jermyn-Mogul Vestry Estate Sevenoaks KentTN145EU Tel: Sevenoaks (0732) 50144 TWX: 95142 Franchised Distributors Lock Distribution Neville Street Chadderton Oldham lancashire Ol9 6lF Tel: Manchester 061-652 0431 TWX: 669619 Sales Representatives International Macro Marketing LId. Burnham lane Slough Berkshire Sl1 6lN Tel: Burnham (062 86) 4422 TWX: 847945 International Argentina Brazil Uruguay Electroimpex SA Guatemala 5991 1425 Buenos Aires, Argentina Tel: 771-3773 Sinchy Rokka's Do Brazil Com E Rep Llda. Rua Cambauba, 6 S/203, Rio De Janeiro, Brazil Tel: (021) 393-1496 Ricagni Importaciones Llda. Av. 18 De Julio, 1216 Montevideo, Uruguai Tel: 90-3671 Transcontinental Marketing Llda. R. Correa Vasques, 58 Sao Paulo, Brazil Tel: (011) 71-3607 6-11 A Schlumberger Company Sales Offices International Austria Fairchild Electronics GmbH Meidlinger Hauptstr. 46 1120Wien Tel: (0043) 0222/858682 TWX: 075096 Japan Pol a Shibuya Bldg., 15-21, Shibuya l-Chome, Shibuya-Ku, Tokyo 150, Japan Tel: (03) 400-8351 Scandinavia Fairchild Semiconductor AB Svartensgatan 6 S-11620 Stockholm, Sweden Tel: 46/8/449255 TWX: 854-17759 Brazil Fairchild Semicondutores Ltda. R. Alagoas, 663 01242 Sao Paulo, Brazil Tel: (011) 66-9092 (011) 67-3224 Yotsubashi Chuo Bldg., 4-26, Shinmachi l-Chome, Nishi-Ku, Osaka 550, Japan Tel: (06) 541-6138 Switzerland Fairchild Camera & Instrument GmbH Baumackerstr. 4 8050 Zurich Tel: (0041) 01/3114230 TWX: 0045-58311 I=AIRCHIL.C France Fairchild 121 Avenue D'italie 75013 Paris, France Tel: 584-55-66 Germany Fairchild Camera & Instrument (Deutschland) GmbH 8046 Garching Daimlerstr. 15 Tel: (0049) 089/320031 TWX: 0524831 Italy Fairchild Semiconduttori S.P.A. Viale Corsica, 7 - 20133 Milano Tel: (02) 296001/5 - 2367741/5 Telegr: Fairsemco TWX: 330522 Fair I Korea Fairchild Semiconductor (Korea) Ltd. No. 219-6, Gari Bong-Dong, Guro-Ku, Seoul, Korea, Tel: 855-0067, 6751 Mexico Blvb. Pte. Adolfo Lopez Mateos No. 163 Col. Mixcoac Delegacion B. Juarez 03910 Mexico D.F. Tel: 563-54-11 Ext: 152, 153, 154, 155 The Netherlands Fairchild Camera & Instrument GmbH Ruysdaelbaan 35 5613 OX-Eindhoven Tel: (0031) 040/446909 TWX: 51024 Fairchild Semiconduttori S.P.A. Via Francesco Saverio Nitti, 11 - 00191 Roma Tel: (06) 3287548-3282717 TWX: 612046 Fair Rom 6-12 United Kingdom Fairchild Camera and Instrument (UK) Ltd. 230 High Street Potters Bar Hertfordshire En6 5BU, England Tel: Potters Bar (0707) 51111 TWX: 262835 Fairchild Camera and Instrument (UK) Ltd. 17 Victoria Street Craigshill Livingston West Lothian EH54 5BG, Scotland Tel: Livingston (0506) 32891 TWX: 72629 FAIRCHILD A Schlumberger Company Fairchild reserves the nght to make changes in the circuitry or specifications in this book at any time without notice. Fairchild cannot assume responsibility for use of any circuitry described other than circuitry embodied in a Fairchild product a other circuit patent licenses are implied , Printed in U.S.A.l214-12-0002-116/50M
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