2000_TI_Audio_Power_Amplifiers 2000 TI Audio Power Amplifiers

User Manual: 2000_TI_Audio_Power_Amplifiers

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"TEXAS
INSTRUMENTS

Audio Power Amplifiers

2000

Analog and Mixed Signal

==================

I uenerai iniormation
Class-D Audio Power Amplifiers
Class-AS Audio Power Amplifiers
Product Previews
Application Reports
Evaluation Modules
Mechanical Data

-

••

~

-

• • 111.

AUQIO I-'ower Amp'ITlers
Data Book

Literature Number: SLOD004

•
TEXAS
INSTRUMENTS

Printed on Recycled Paper

IMPORTANT NOTICE
Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue
any product or service without notice, and advise customers to obtain the latest version of relevant information
to verify, before plaCing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at .the time of order acknowledgment, including those
pertaining to warranty, patent infringement, and limitation of liability.
TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with TI's standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily
performed, except those mandated by government requirements.
Customers are responsible for their applications using TI components.
In order to minimize risks associated with the customer's applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product deSign. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. TI's publication of information regarding any third
party's products or services does not constitute TI's approval, warranty or endorsement thereof.

Copyright © 2000, Texas Instruments Incorporated

Printed in U.S.A by
Von Hoffmann Graphics, Inc.
Owensville, Missouri

INTRODUCTION

What you will find
inside...

Texas Instruments' Audio Power Amplifier (APA) Data Book presents technical
information on over 40 differentiated APAs from TI. This includes product previews of the soon to be released families.
An entire section on application notes gives insight into how to select APAs and
answers to frequently asked design questions. Following the application notes
section is an overview of all TI's APA design tools.
The Plug-n-Play EVMs and software tools are developed with one goal in mind:
Minimize Design Time. The final section contains packaging specifications, including tape and reel dimensions for the ultra-small MSOP PowerPADTM package.

How the data book
is organized ...

New products and
applications ...

1) Introduction and general information
2) Class-D APA Datasheets (sorted ascending by output power)
3) Class-AS APA Datasheets (sorted ascending by output power)
4) Preliminary Class-D and Class-AS Datasheets
5) Application Notes (sorted alphabetically by title)
6) Design Tools
7) Mechanical Data

•
•
•
•
•
•

TPA032DOx
TPA2000D2
TPA01x2
TPA02x2
TPA02x3
TPA0211

•
•
•
•
•

Notebook PCs
Multimedia Speakers
Wireless Speakers
Hands-Free Car Kits
P.O.S. Terminals

•
•
•

TPA7x1
TPA3x1
TPA1x2

•
•
•

Wireless Phones
InterneVPersonal Audio
Personal FM Transceivers

Where to go for
Download TI's latest datasheets and applications notes via the internet at:
more information ... http:Uwww.ti.comfscfdocsfschome.htm To provide full technical support, Texas
Instruments has a large fully-staffed technical information center available to help
you. Please turn to the last page of this data book for a complete listing of contacts ready to answer your questions.

v

vi

I Generai information

1-1

Contents
Page
Alphanumeric Index ............................................ . . . . . . . . .. 1-3
How To Select an Audio Power Amplifier ............................... 1-5
Cross Reference ......................................................... 1-14
Glossary .................................................................. 1-18

C)
CD

:s

CD

-....o
iiJ

:s
~

3
m
...._.
o
:s

1-2

ALPHANUMERIC INDEX

TPA005D02
TPA005D12
TPA005D14
TPA0102
TPA0103
TPA0112
TPA0122
TPA0132
TPA0142
TPA0152
TPA0162
TPA0202
TPA0211
TPA0212
TPA0213
TPA0222
TPA0223
TPA0232
TPA0233
TPA0242
TPA0243
TPA0253
TPA032D01
TPA032D02
TPA032D03
TPA032D04
TPA102
TPA112
TPA122
TPA152
TPA301
TPA302
TPA311
TPA701
TPA711
TPA721
TPA1517
TPA2000D2
TPA4860
TPA4861

2-W Class-D Stereo Audio Power Amplifier .................................... 2-53
2-W Class-D Stereo Audio Power Amplifier .................................... 2-19
2-W Class-D Stereo Audio Power Amplifier .................................... 2-25
2-W Stereo Audio Power Amplifier ........................................... 3-313
1.75-W Three-Channel Audio Power Amplifier ................................. 3-277
2-W Stereo Audio Power Amplifier ........................................... 3-349
2-W Stereo Audio Power Amplifier ........................................... 3-381
2-W Stereo Audio Power Amplifier ........................................... 3-413
2-W Stereo Audio Power Amplifier ........................................... 3-441
2-W Stereo Audio Power Amplifier ........................................... 3-469
2-W Stereo Audio Power Amplifier ........................................... 3-497
2-W Stereo Audio Power Amplifier ........................................... 3-525
2-W Mono Audio Power Amplifier .............................................. 4-3
2-W Stereo Audio Power Amplifier ........................................... 3-565
2-W Mono Audio Power Amplifier ............................................ 3-597
2-W Stereo Audio Power Amplifier ........................................... 3-607
2-W Mono Audio Power Amplifier ............................................ 3-639
2-W Stereo Audio Power Amplifier ........................................... 3-643
2-W Mono Audio Power Amplifier ............................................ 3-671
2-W Stereo Audio Power Amplifier ........................................... 3-675
2-W Mono Audio Power Amplifier ............................................ 3-703
1-W Mono Audio Power Amplifier ............................................ 3-271
10-W Class-D Mono Audio Power Amplifier .................................... 2-77
10-W Class-D Stereo Audio Power Amplifier ................................... 2-97
10-W Class-D Mono Audio Power Amplifier ................................... 2-119
10-W Class-D Stereo Audio Power Amplifier .................................. 2-141
150-mW Stereo Audio Power Amplifier ........................................ 3-17
150-mW Stereo Audio Power Amplifier ........................................ 3-39
150-mW Stereo Audio Power Amplifier ........................................ 3-63
75-mW Stereo Audio Power Amplifier .......................................... 3-3
350-mW Stereo Audio Power Amplifier ....................................... 3-105
300-mW Stereo Audio Power Amplifier ........................................ 3-85
350-mW Stereo Audio Power Amplifier ....................................... 3-125
700-mW Stereo Audio Power Amplifier ....................................... 3-155
700-mW Stereo Audio Power Amplifier ....................................... 3-175
700-mW Stereo Audio Power Amplifier ....................................... 3-205
6-W Stereo Audio Power Amplifier ........................................... 3-707
2-W Filterless Stereo Class-D Audio Power Amplifier ............................. 2-3
1-W Stereo Audio Power Amplifier ........................................... 3-225
1-W Stereo Audio Power Amplifier ........................................... 3-249

The devices in BOLD type are in the Product Preview stage of development.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

1-3

ALPHANUMERIC INDEX

Part Number

PsgeNumber

Part Description

Class·D Audio Power Amplifiers
TPA2000D2

2-W Filterless Stereo Class-D Audio Power Amplifier

2-3

TPA005D12

2-W Class-D Stereo Audio Power Amplifier

2-19

TPAOO5D14

2-W Class-D Stereo Audio Power Amplifier

2-25

TPAOO5D02

2-W Class-D Stereo Audio Power Amplifier

2-53

TPA032D01

10-W Class-D Mono Audio Power Amplifier

2-77

TPA032D02

10-W Class-D Stereo Audio Power Amplifier

2-97

TPA032D03

1Q-W Class-D Stereo Audio Power Amplifier

2-119

TPA032D04

10-W Class-D Stereo Audio Power Amplifier

2-141

Class-AB Audio Power Amplifiers
TPA152

75-mW Stereo Audio Power Amplifier

3-3

TPA102

15Q-mW Stereo Audio Power Amplifier

3-17

TPA112

15Q-mW Stereo Audio Power Amplifier

3-39

TPA122

15Q-mW Stereo Audio Power Amplifier

3-63

TPA302

300-mW Stereo Audio Power Amplifier

3-85

TPA301

350-mW Stereo Audio Power Amplifier

3-105

TPA311

350-mW Stereo AudiO Power Amplifier

3-125

TPA701.

700-mW Stereo Audio Power Amplifier

3-155

TPA711

70Q-mW Stereo Audio Power Amplifier

3-175

TPA721

700-mW Stereo Audio Power Amplifier

3-205

TPA4860

1-W Stereo Audio Power Amplifier

3-225

TPA4861

1-W Stereo Audio Power Amplifier

3-249

TPA0253

1-W Mono Audio Power Amplifier

3-271

TPA0103

1.75 Three-Channel Audio Power Amplifier

3-277

TPA0102

2-W Stereo Audio Power Amplifier

3-313

TPA0112

2-W Stereo Audio Power Amplifier

3-349

TPA0122

2-W Stereo Audio Power Amplifier

3-381

TPA0132

2-W Stereo Audio Power Amplifier

3-413

TPA0142

2-W Stereo Audio Power Amplifier

3-441

TPA0152

2-W Stereo Audio Power Amplifier

3-469

TPA0162

2-W Stereo Audio Power Amplifier

3-497

TPA0202

2-W Stereo Audio Power Amplifier

3-525

TPA0212

2-W Stereo Audio Power Amplifier

3-565

TPA0213

2-W Mono Audio Power Amplifier

3-597

TPA0222

2-W Stereo Audio Power Amplifier

3-607

TPA0223

2-W Mono Audio Power Amplifier

3-639

TPA0232

2-W Stereo Audio Power Amplifier

3-843

TPA0233

2-W Mono Audio Power Amplifier

3-671

TPA0242

2-W Stereo Audio Power Amplifier

3-675

TPA0243

2-W Mono Audio Power Amplifier

3-703

TPA1517

6-W Stereo Audio Power Amplifier

3-707

Preliminary Datasheets
TPA0211

I

2-W Mono Audio Power Amplifier

~'TEXAS

INSTRUMENTS

1-4

POST OFFICE eox 655303 • DAUAS. TEXAS 75265

4-3

HOW TO SELECT AN AUDIO POWER AMPLIFIER

How to Select an Audio Power Amplifier
Introduction
This section is written to help guide designers that are needing an audio power amplifier in a new or existing
design. TI's large portfolio of over 35 devices provides a designer many options to choose from and helps insure
a near perfect fit in their application. However, the quantity of products makes choosing the correct audio
amplifier more difficult and time consuming. Knowing what devices map to which applications and the
differentiating specifications that are most important help minimize the effort and time in the selection process.
Table 1 maps Tl's current offering of APAs to end equipment.

Table 1. End Equipment With Suggested TI APA Solution
Wireless Phones and Personal FM Transceivers
Key Features

Device
TPA701

TPA711

TPA721

TPA0211t

•
•
•
•
•
•
•
•
•
•
•
•
•

TPA102

TPA112

TPA122

TPA152

TPA301

TPA311

3-155

700-mW mono speaker output drive
Configured to drive both speakers and headphones
Tiny 8-pin MSOP PowerPAD package reduces PCB size

3-175

700-mW mono speaker output drive
Differential input for improved CMR
Tiny 8-pin MSOP PowerPAD package reduces PCB size

3-205

2-W mono speaker output drive
Configured to drive both speakers and headphones
Tiny 8-pin MSOP PowerPAD package reduces PCB size
Upgrade to the TPA711 and TPA4861

4-3

Internet and Personal Audio
Key Features

Device

•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•

Page

700-mW mono speaker output drive
Ultra-low shutdown control maximizes battery life
Tiny 8-pin MSOP PowerPAD package reduces PCB size

Page

150-mW output into stereo headphones
Shutdown control for maximum battery life
Tiny 8-pin MSOP PowerPAD package reduces PCB size

3-17

150-mW output into stereo headphones
Shutdown control for maximum battery life
Tiny 8-pin MSOP PowerPAD package reduces PCB size

3-39

150-mW output into stereo headphones
Shutdown control for maximum battery life
Tiny 8-pin MSOP PowerPAD package reduces PCB size

3-63

Hi-Fi 75-mW stereo headphone driver
Improved depop circuitry

3-3

350-mW mono speaker output drive
Low supply current and shutdown current for long battery life
Tiny 8-pin MSOP PowerPAD package reduces PCB size

3-105

350-mW mono speaker output drive
Configured to drive both speakers and headphones
Tiny 8-pin MSOP PowerPAD package reduces PCB size

3-125

This device is in the Product Preview sta9e of develo pmen!. Contact you local TI sales office for more information.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

1-5

HOW TO SELECT AN AUDIO POWER AMPLIFIER

Device
TPA701

TPA711
TPA721

TPA0211t

TPA0213

TPA0223

TPA0233

•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
••
•
•
•
•
•
•
•

TPA0243

••
•

Internet and Personal Audio (continued)
Key Features
700-mW mono speaker output drive
Ultra low shutdown control maximizes battery life
Tiny 8-pin MSOP PowerPAD package reduces PCB size
700-mW mono speaker output drive
Configured to drive both speakers and headphones
Tiny 8-pin MSOP PowerPAD package reduces PCB size
700-mW mono speaker output drive
Tiny 8-pin MSOP PowerPAD package reduces PCB size
2-W mono speaker output drive
Configured to drive both speakers and headphones
Tiny 8-pin MSOP PowerPAD package reduces PCB size
Upgl'lilde to the TPA711 and TPA4861
2-W mono speaker output drive
Separate mono and stereo inputs for maximum flexibility
Optimized for battery life
Stereo headphone drive
Tiny 10-pin MSOP PowerPAD package reduces PCB size
2-W mono speaker output drive
Separate mono and stereo inputs for maximum flexibility
Optimized for fidelity
Stereo headphone drive
Tiny 10-pin MSOP PowerPAD package reduces PCB size
2-W mono speaker output drive
Mono output generated from internally mixed stereo inputs reduce external
components
Optimized for battery life
Stereo headphone drive
Tiny 10-pin MSOP PowerPAD package reduces PCB size
2-W mono speaker output drive
Mono output generated from internally mixed stereo inputs reduce external
components
Optimized for fidelity
Stereo headphone drive
Tiny 10-pin MSOP PowerPAD package reduces PCB size
1-W mono speaker output drive
Stereo headphone drive
Ultra low supply current and shutdown current for maximum battery life
Tiny 10-pin MSOP PowerPAD package reduces PCB size

•
•
TPA0253
•
•
This device Is in the Product Preview sta e of developmen!. Contact ou local TI sales office for more information.
g

1-6

y

:lllExAs
INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

Page
3-155

3-175
3-205

4-3

3-597

3-639

3-671

3-703

3-271

HOW TO SELECT AN AUDIO POWER AMPLIFIER

Notebook PC

Device

Key Features

•
•
•
Industry standard 2-W stereo speaker output drive
TPAOO5D12 •
• Efficient Class-D operation generates minimal heat and extends battery life
• Industry standard 2-W stereo speaker output drive
TPAOO5D14 • Efficient Class-D operation generates minimal heat and extends battery life
• Stereo headphone drive
standard 2-W stereo speaker output drive
• Industry
Internal gain settings reduce external components
•
TPA0112
• Stereo headphone drive
• Optimized for battery life
standard 2-W stereo speaker output drive
• Industry
Internal gain settings reduce external components
• Stereo
TPA0122
headphone drive
• Optimized
for fidelity
•
Industry standard 2-W stereo speaker output drive
• DC
volume control increases flexibility and reduces external components
• Stereo
TPA0132
headphone drive
• Optimized
for battery life
•
Industry
standard
stereo speaker output drive
• DC volume control2-W
increases flexibility and reduces external components
•
TPA0142
Stereo headphone drive
• Optimized
for fidelity
•
Industry standard 2-W stereo speaker output drive
• Digital
control increases flexibility and reduces external components
• Stereo volume
TPA0152
headphone drive
• Optimized
for battery life
•
Industry
standard
2-W stereo speaker output drive
• Digital volume control
increases flexibility and reduces external components
•
TPA0162
Stereo headphone drive
• Optimized
for fidelity
•
Industry standard 2-W stereo speaker output drive
•
TPA0202
• Industry's lowest THD+N provides hi-fi performance
• Stereo headphone drive
• 1.5-W stereo speaker output drive
TPA0102
• Stereo headphone drive
standard 2-W stereo speaker output drive
• Industry
Internal gain settings reduce external components
• Separate
• (SE/BTL) input MUX control pin for maximum control of the amplifier configuration
TPA0212
Stereo headphone drive
• Optimized
for battery life
•
This device is in the Product Preview sta e of develo pmen!. Contact ou local TI sales office tor more information.
TPA2000D2

No output filter required
Efficient Class-D operation generates minimal heat and extends battery life
Industry standard 2-W stereo speaker output drive

9

Page
2-3

2-21

2-27

3-39

3-63

3-413

3-441

3-3

3-497

3-525

3-17

3-565

y

~TEXAS

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1-7

HOW TO SELECT AN AUDIO POWER AMPLIFIER

Notebook PC (continued)
Key Features

Device

TPA0222

TPA0232

•
•
•
•
•
•
•
•

•
•
•

TPA0242

•
•
•

•

TPA0213

TPA0223

TPA0233

•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•

Page

Industry standard 2-W stereo speaker output drive
Intemal gain settings reduce extemal components
Separate input MUX control pin for maximum control of the amplifier configuration
(SE/BTL)
Stereo headphone drive
Optimized for fidelity

3-607

Industry standard 2-W stereo speaker output drive
DC volume control increases flexibility and reduces extemal components
Separate input MUX control pin for maximum control of the amplifier configuration
(SE/BTL)
Stereo headphone drive
Optimized for battery life

3-643

Industry standard 2-W stereo speaker output drive
DC volume control increases flexibility and reduces extemal components
Separate input MUX control pin for maximum control of the amplifier configuration
(SE/BTL)
Stereo headphone drive
Optimized for fidelity

3-675

Industry standard 2-W mono speaker output drive
Separate mono and stereo inputs for maximum flexibility
PC 99 Compatible (Portable)
Optimized for battery life
Stereo headphone drive

3-597

Industry standard 2-W mono speaker output drive
Separate mono and stereo inputs for maximum flexibility
PC 99 Compatible (Desktop)
Optimized for fidelity
Stereo headphone drive

3-639

Industry standard 2-W mono speaker output drive
Mono output generated from intemally mixed stereo inputs to reduce extemal
components
PC 99 Compatible (Portable)
Optimized for battery life
Stereo headphone drive

3-671

Industry standard 2-W mono speaker output drive
Mono output generated from intemally mixed stereo inputs to reduce extemal
components
TPA0243
PC 99 Compatible (Desktop)
Optimized for fidelity
Stereo headphone drive
This device is in the Product Preview sta9e of developmen!. Contact you local TI sales office for more information.

•

•
•
•

~TEXAS

INSTRUMENTS
1-8

POST OFFICE BOX 65S303 • DALlAS. TEXAS 75265

3-703

HOW TO SELECT AN AUDIO POWER AMPLIFIER

Multimedia and Wireless Speakers
Key Features

Device

•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•
•

TPA2000D2

TPAOO5D12

TPAOO5D14

TPA032D01
TPA032D02

TPA032D03

TPA032D04

Page

No output filter required
Efficient Class-D operation generates minimal heat and extends battery life
Industry standard 2-W stereo speaker output drive

2-3

2-W stereo output drive for satellite speakers
Efficient Class-D operation generates minimal heat and extends battery life

2-21

2-W stereo output drive for satellite speakers
Efficient Class-D operation generates minimal heat and extends battery life
Stereo headphone drive

2-27

10-W mono output drive for sub-woofer or satellite speakers
Efficient Class-D operation generates minimal heat eliminating bulky heat sinks

2-79

10-W stereo output drive for sub-woofer or satellite speakers
Efficient Class-D operation generates minimal heat eliminating bulky heat sinks

2-99

10-W stereo output drive for sub-woofer or satellite speakers
Efficient Class-D operation generates minimal heat eliminating bulky heat sinks
Stereo headphone drive

2-121

10-W stereo output drive for sub-woofer or satellite speakers
Efficient Class-D operation generates minimal heat eliminating bulky heat sinks
Stereo headphone drive

2-143

Determining Output Power When Driving Headphones (Single Ended) vs. Speakers (Bridged)
The configuration of the amplifier dramatically affects how much power can be delivered to the speaker. Single
ended (SE) configuration is most common in headphone or applications when the speakers use a common
ground. It is referred to as single ended because only one terminal of the speaker is connected to the amplifier.
The other terminal is tied to ground, see Figure 1. This technique requires only three conductors between the
amplifier and speaker for a stereo solution, left positive, right positive and the third for ground. In terms of power
provided to the load, the equation is straight forward, just remember to convert the supply voltage to an RMS
value by dividing the peak to peak voltage by 2 x (2)1/2 or 2.83. Once VRMS is determined plug the value into
Equation 1 to find the power delivered to the speaker:

(1 )

p

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

1-9

HOW TO SELECT AN AUDIO POWER AMPLIFIER

325 kf.l
RF

.:l-

325 kf.l

VOO 6

i

VOot2

VOO

cS

-=-

Audio
Input

~

I~

RI

8

·V

IN1-

V01 7

I-

I

+

CI
1

BYPASS

4

IN 2-

ICC

CBl-

Audio
Input

~C

RI

~

T
7

V02 5

I+

I

FromShutdown
Control Clrcult

3

SHUTOOWN

I
I

1.
ICC

Bias
Control

I Il
2

-

RF

Figure 1. TPA 102 Audio Power Amplifier in SE Configuration
A bridge-tied load (BTL) configuration consists of two amplifiers driving both ends of the load, see Figure 2.
There are several potential benefits with this configuration. The first benefit is the elimination of the coupling
capacitor requirement in the SE configuration used to block the DC offset from reaching the load. These
capacitors can be quite large (40 -1000 uF), are expensive and have the additional drawback of limiting low
frequency performance. The BTL configuration cancels the DC offsets which eliminates the need for the
blocking caps. Low frequency performance is then limited only by the input network, amplifier and speaker
frequency response. The other major advantage is the differential drive to the speaker. The differential drive
means that as one side is slewing up the other side is slewing down and vice versa. This effectively doubles
the available voltage swing on the load. Doubling the voltage swing across the speaker quadruples the power
delivered to the speakers.
BTL configurations are typically used in applications when the speaker and amplifier are contained in the same
enclosure. For example, the circuit in Figure 2 is useful in wireless applications where only a mono speaker is
required. The APA is capable of driving 700 mW to an a-ohm speaker from a 5-V supply.

-!I1TEXAS
1-10

INSTRUMENTS

POST OFFICE BOX $5303 • DALLAS. TEXAS 75265

HOW TO SELECT AN AUDIO POWER AMPLIFIER

VDD 6
RF

J

Audio
Input

~c

RI

~

I

4

IN-

3

IN+

2

BYPASS

VDD/2

r

,
,
,
,
,
,
,
,
,
,
,
,

CBT

-=-

1

Fro m System Control

SHUTDOWN

Cs

~

VO+ 5

---. Y
r

I Bias
I Control

-

•

I

±

VDD

I
1

~

y

a=r(
..........

Vo-

-=-

700mW

7
GND

J-

Figure 2. TPA701 Audio Power Amplifier in BTL Configuration

Determining the correct supply voltage to avoid clipping
The output voltage swing is key when determining the peak power capability of an amplifier. Figure 3 and Figure
4 show the theoretical output power from a 5-V supply into a 4-ohm load is 781 mW (SE) and 3.12 W (BTL)
respectively. However, to avoid clipping the APA output voltage should not swing rail-to-rail. A few tenths to a
volt of headroom from the top supply rail significantly decreases distortion (clipping).
For example, an amplifier with a 5-V single supply, driving a 4-ohm speaker has a typical peak to peak output
swing around 4.5 V. This translates into 1.59 VRMS If the speakers are 4 ohms and the supply voltage is 5 V the
maximum output power from a SE and BTL configuration is:
( 4.5 V)2

( 9 V )2

40

40

2.83V

P SE

=

2.83V

P BTL = 2.53 W

0.63 W

A resultfrom this analysis is lower speaker impedance yields higher output power. However, speakers with lower
impedance are typically less efficient, especially speakers with an impedance below 4 ohms. Moreover, the
APfJ\s efficiency decreases as the speaker's impedance drops below 4 ohms. The degradation in the speaker's
and APfJ\s efficiency below 4 ohms negates the increase in output power.
Beyond lowering the speaker impedance to 4 ohms, the best way to increase the output power in a given SE
or BTL configuration is by increasing the supply voltage. Figure 3 and Figure 4 are plots of the maximum
theoretical output power vs supply voltage for SE and BTL amplifier configurations driving 4-, 8- and 32-ohm
speakers.

~TEXAS

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1-11

HOW TO SELECT AN AUDIO POWER AMPLIFIER

SINGLE·ENDED CONFIGURATION
MAXIMUM THEORECTICAL OUTPUT POWER
VB
SUPPLY VOLTAGE
8
7

/

6
~

RL=40/

I

J
I
I

2

5

I

4
3

2

(0.78W>/

o
o

~
2.5

5

V RL=8~~'
,p/ .,"
-~ " RL=.~Oii
~

7.5

10

12.5

15

VDD - Supply Voltage - V

Figure 3. Maximum Theoretical Output Power vs Supply Voltage for a SE Audio Power Amplifier
BRIDGE·TlED LOAD CONFIGURATION
MAXIMUM THEORECTICAL OUTPUT POWER
VB
SUPPLY VOLTAGE
30

I

25
~
I

J

I
I

RL=40/
20

/

15
10

2

V

--'
h-; ~- ""'"

5 r - - (3.12W) /

o
o

V RL=80/'
,

/

2.5

5

7.5

~"

"

--

-

RL=320.-,
~.

10

12.5

15

VDD - Supply Voltage - V

Figure 4. Maximum Theoretical Output Power vs Supply Voltage for a BTL Audio Power Amplifier

~1ExAs

1-12

INSTRUMENTS
POST OFFICE BOX 655303 • DAll.AS, TEXAS 75265

HOW TO SELECT AN AUDIO POWER AMPLIFIER

Conclusion
Knowing the maximum output power a given APA can deliver from a fixed supply voltage will save considerable
time and effort when selecting a device. An APA with a BTL configuration will drive four times more power to
the speaker than an APA in a SE configuration. Once the amplifier output configuration is selected there are
basically two variables that limit the output power being supplied to the speaker; the APNs supply voltage and
the speaker's impedance. Lowering the impedance of the speaker will increase the APNs output power, but the
loss in speaker efficiency tends to offset the increase in output power. This means the only way to effectively
increase the output power from a speaker is to increase the supply voltage to the amplifier.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

1-13

AUDIO POWER AMPLIFIER CROSS REFERENCE

Part No.

Suggested TI
Replacement

Vendor

Replacement Type

Page No.

LX1720

TPA2000D2

LinFinity

Similar functionality (see Note 4)

2-3

LX1720

TPA032D02

LinFinity

Similar functionality (see Note 4)

2-97

MS6308

TPA102

MOSA

Same functionality (see Note 3)

3-17

MS6308

TPAl12

MOSA

Same functionality and pinout (see Note 2)

3-39

MS6308

TPA122

MOSA

Same functionality (see Note 3)

3-63

MS6308

TPA152

MOSA

Same functionality (see Note 3)

3-3

SSM2211

TPA4861

Analog Devices

Same functionality (see Note 3)

3-249

SSM2211

TPA701

Analog Devices

Same functionality (see Note 3)

3-155

SSM2211

TPA0211t

Analog Devices

Similar functionality (see Note 4)

4-3

SSM2250

TPA0213

Analog Devices

Similar functionality (see Note 4)

3-597

SSM2250

TPA0223

Analog Devices

Similar functionality (see Note 4)

3-639

SSM2250

TPA0233

Analog Devices

Similar functionality (see Note 4)

3-671

SSM2250

TPA0243

Analog Devices

Similar functionality (see Note 4)

3-703

MC34119

TPA301

Motorola

Same functionality (see Note 3)

3-105

MC34119

TPA701

Motorola

Similar functionality (see Note 4)

3-155

TDA8542

TPA0102

Philips

Similar functionality (see Note 4)

3-313

TDA8542

TPA0202

Philips

Similar functionality (see Note 4)

3-525

TDA8542

TPAOl12

Philips

Similar functionality (see Note 4)

3-349

TDA8542

TPA0122

Philips

Similar functionality (see Note 4)

3-381

TDA8542

TPA0212

Philips

Similar functionality (see Note 4)

3-565

TDA8542

TPA0222

Philips

Similar functionality (see Note 4)

3-607

TDA7053A

TPA0132

Philips

Similar functionality (see Note 4)

3-413

TDA7053A

TPA0142

Philips

Similar functionality (see Note 4)

3-441

TDA7053A

TPA0232

Philips

Similar functionality (see Note 4)

3-643

TDA7053A

TPA0242

Philips

Similar functionality (see Note 4)

3-675

TDA1308

TPA152

Philips

Same functionality (see Note 3)

3-3

TDA1308

TPA102

Philips

Same functionality (see Note 3)

3-17

TDA1308

TPAl12

Philips

Same functionality (see Note 3)

3-39

TDA1308

TPA122

Philips

Same functionality (see Note 3)

3-63

t This device is in the Product Preview stage of development. Contact your local TI sales office for more information.
NOTES:

1.
2.
3.
4.

The device is an EXACT EQUIVALENT in functionality and parametrics to the competitors device.
The device has the Same functionality and pinout as the compet~ors device, but Is NOT and exact equivalent
The device has the Same functionality as the compemors device, but is not pin-for-pin and/or parametrically equivalent.
The device has Similar functionality. but is not functionally equivalent to the competitors device.

~TEXAS

INSTRUMENTS
1-14

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

AUDIO POWER AMPLIFIER CROSS REFERENCE

Part No.

Suggested TI
Replacement

Vendor

Replacement Type

Page No.

TDA8559

TPA152

Philips

Same functionality (see Note 3)

3-3

TDA8559

TPA102

Philips

Same functionality (see Note 3)

3-17

TDA8559

TPA112

Philips

Same functionality (see Note 3)

3-39

TDA8559

TPA122

Philips

Same functionality (see Note 3)

3-63

TDA1517

TPA1517

Philips

Same functionality (see Note 3)

3-707

TDA7052

TPA4861

Philips

Same functionality (see Note 3)

3-249

TDA7052

TPA0211t

Philips

Same functionality (see Note 3)

4-3

TDA7052A

TPA0132

Philips

Similar functionality (see Note 4)

3-413

TDA7052A

TPA0142

Philips

Similar functionality (see Note 4)

3-441

TDA7052A

TPA0232

Philips

Similar functionality (see Note 4)

3-643

TDA7052A

TPA0242

Philips

Similar functionality (see Note 4)

3-675

TDA8552

TPA0152

Philips

Similar functionality (see Note 4)

3-469

TDA8552

TPA0162

Philips

Similar functionality (see Note 4)

3-497

TDA8551

TPA0152

Philips

Similar functionality (see Note 4)

3-469

TDA8551

TPA0162

Philips

Similar functionality (see Note 4)

3-497

LM4663

TPA2000D2

National Semiconductor

Similar functionality (see Note 4)

2-3

LM4663

TPA005D14

National Semiconductor

Similar functionality (see Note 4)

2-25

LM4862

TPA701

National Semiconductor

Same functionality (see Note 3)

3-155

LM4862

TPA711

National Semiconductor

Similar functionality (see Note 4)

3-175

LM4862

TPA721

National Semiconductor

Similar functionality (see Note 4)

3-205

LM4862

TPA301

National Semiconductor

Similar functionality (see Note 4)

3-105

LM4862

TPA311

National Semiconductor

Similar functionality (see Note 4)

3-125

LM4835

TPA0132

National Semiconductor

Similar functionality (see Note 4)

3-413

LM4835

TPA0142

National Semiconductor

Similar functionality (see Note 4)

3-441

LM4835

TPA0232

National Semiconductor

Similar functionality (see Note 4)

3-643

LM4835

TPA0242

National Semiconductor

Similar functionality (see Note 4)

3-675

LM4835

TPA0112

National Semiconductor

Similar functionality (see Note 4)

3-349

LM4835

TPA0122

National Semiconductor

Similar functionality (see Note 4)

3-381

LM4835

TPA0212

National Semiconductor

Similar functionality (see Note 4)

3-565

t This device is in the Product Preview stage of development. Contact your local TI sales office for more information.
NOTE:'>:

1.
2.
3.
4.

The device is an EXACT EQUIVALENT in functionality and parametrics to the compeiHors device.
The device has the Same functionality and pinout as the competitors device, but is NOT and exact equivalent
The device has the Same functionality as the competitors device, but is not pin-for-pin and/or parametrically equivalent.
The device has Similar functionality, but is not functionally equivalent to the competitors device.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

1-15

AUDIO POWER AMPLIFIER CROSS REFERENCE

Part No.

Suggested TI
Replacement

Vendor

Replacement Type

Page No.

LM4835

TPA0222

National Semiconductor

Similar functionality (see Note 4)

3-607

LM386

TPA301

National Semiconductor

Similar functionality (see Note 4)

3-105

LM4865

TPA711

National Semiconductor

Similar functionality (see Note 4)

3-175

LM4865

TPA0132

National Semiconductor

Similar functionality (see Note 4)

3-413

LM4865

TPA0142

National Semiconductor

Similar functionality (see Note 4)

3-441

LM4865

TPA0232

National Semiconductor

Similar functionality (see Note 4)

3-643

LM4865

TPA0242

National Semiconductor

SimHar functionality (see Note 4)

3-675

LM4752

TPA032D02

National Semiconductor

Similar functionality (see Note 4)

2-97

LM4880

TPA122

National Semiconductor

Same functionality and pinout (see Note 2)

3-63

LM4880

TPA102

National Semiconductor

Same functionality (see Note 3)

3-17

LM4880

TPA112

National Semiconductor

Same functionality (see Note 3)

3-39

LM4881

TPA102

National Semiconductor

Same functionality and pinout (see Note 2)

3-17

LM4881

TPA112

National Semiconductor

Same functionality (see Note 3)

3-39

LM4881

TPA122

National Semiconductor

Same functionality and pinout (see Note 2)

3-63

LM4882

TPA311

National Semiconductor

Similar functionality (see Note 4)

3-125

LM4882

TPA301

National Semiconductor

Similar functionality (see Note 4)

3-105

LM4871

TPA4861

National Semiconductor

Same functionality and pinout (see Note 2)

3-249

LM4871

TPA701

National Semiconductor

Same functionality and pinout (see Note 2)

3-155

LM4871

TPA0211t

National Semiconductor

Similar functionality (see Note 4)

4-3

LM4864

TPA301

National Semiconductor

Same functionality and pinout (see Note 2)

3-105

LM4864

TPA311

National Semiconductor

Similar functionality (see Note 4)

3-125

LM4873

TPA0102

National Semiconductor

Same functionality (see Note 3)

3-313

LM4873

TPA0202

National Semiconductor

Same functionality (see Note 3)

3-525

LM4873

TPA0112

National Semiconductor

Similar functionality (see Note 4)

3-349

LM4873

TPA0122

National Semiconductor

Similar functionality (see Note 4)

3-381

LM4873

TPA0212

National Semiconductor

Similar functionality (see Note 4)

3-565

LM4873

TPA0222

National Semiconductor

Similar functionality (see Note 4)

3-607

LM4863

TPA0102

National Semiconductor

Similar functionality (see Note 4)

3-313

LM4863

TPA0202

National Semiconductor

Similar functionality (see Note 4)

3-525

t This device is in the Product Preview stage of development. Contact your local TI sales office for more information.
NOTES: 1. The device is an EXACT EQUIVALENT in functionality and parametrics to the competitors device.
2. The device has the Same functionality and pinout as the competitors device, but is NOT and exact equivalent
3. The device has the Same functionality as the competitors device, but is not pin-for-pin and/or parametrically equivalent.
4. The device has Similar functionality, but is not functionally equivalent to the competitors device.

~TEXAS

1-16

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

AUDIO POWER AMPLIFIER CROSS REFERENCE

Part No.

Suggested Tl
Replacement

Vendor

Replacement Type

Page No.

LM4863

TPAOl12

National Semiconductor

Similar functionality (see Note 4)

3-349

LM4863

TPA0122

National Semiconductor

Similar functionality (see Note 4)

3-381

LM4863

TPA0212

National Semiconductor

Similar functionality (see Note 4)

3-565

LM4863

TPA0222

National Semiconductor

Similar functionality (see Note 4)

3-607

LM4861

TPA4861

National Semiconductor

Same functionality and pinout (see Note 2)

3-249

LM4861

TPA0211t

National Semiconductor

Similar functionality (see Note 4)

LM4860

TPA4860

National Semiconductor

Same functionality and pinout (see Note 2)

3-225

LM4834

TPA0132

National Semiconductor

Similar functionality (see Note 4)

3-413

LM4834

TPA0142

National Semiconductor

Similar functionality (see Note 4)

3-441

LM4834

TPA0232

National Semiconductor

Similar functionality (see Note 4)

3-643

LM4834

TPA0242

National Semiconductor

Similar functionality (see Note 4)

3-675

t This device
NOTES: 1.
2.
3.
4.

4-3

is in the Product Preview stage of development. Contact your local TI sales office for more information.
The device is an EXACT EQUIVALENT in functionality and parametrics to the competitors device.
The device has the Same functionality and pinout as the competitors device, but is NOT and exact equivalent
The device has the Same functionality as the competitors device, but is not pin-for-pin and/or parametrically equivalent.
The device has Similar functionality, but is not functionally equivalent to the competitors device.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

1-17

AUDIO POWER AMPLIFIER
GLOSSARY
Single-Ended Load Configuration

A configuration where one end of the load is connected to the audio power amplifier and the other end of the
load is connected to ground. Used primary for headphone applications or where the audio power amplifier and
speaker reside in different enclosures.
Bridged-Tied Load Configuration

A configuration where both ends of the load are connected to audio power amplifiers. This configuration
effectively quadruples the output power capability of the system. Used primary in applications that are space
constrained and where the audio power amplifier and speaker reside in the same enclosure.
PWM (Pulse Width Modulation)

Pulse-time modulation in which the value of each instantaneous sample of the modulating wave is caused to
modulate the duration of a pulse. The modulation frequency may be fixed or variable. PWM is used in Class-D
audio power amplifiers to achieve very high efficiency operation.
Class-A Amplifiers

Class-A, based on one output element, a vacuum tube, which was eventually replaced by a transistor. Class-A
amplifiers add little distortion to the sound they amplify, But, they consume a great deal of power. In many
applications, this would require systems with very large power supplies. As a result, the effective use of Class-A
amplifiers in portable applications is severely limited.
Class-B Amplifiers

Class-B addressed the problem of power consumption. This type of APA features two elements or transistors
in the output stage, both of which are shut off when no signal is present. Unfortunately, this arrangement
introduces significant distortion into the signal as it moves through the zero crossover point.
Class-AB Amplifiers

Class-AB amplifiers removed the distortion by keeping each of the two transistors slightly on at all times. While
this improves THD+N it also re-introduces the problem of power consumption. Class-AB amplifiers are ideal
solutions in applications requiring moderate to high levels of fidelity and supply current.
Class-D Amplifiers

Class-D amplifiers process analog Signals using PWM techniques, which is the key behind Class-D amplifiers'
increased efficiency. The PWM signals are applied to power DMOS H-bridges, which provide high output current
capability. High-frequency square waves of constant amplitude, but varying width, are output from the IC. These
pulses of varying widths contain the audio information.
Total Harmonic Distortion + Noise (THD+N)

The root some square of all harmonic distortion components including their aliases plus any noise in the system.
Commonly measured as a percentage ofthe fundamental Signal. Harmonic distortion is distortion at frequencies
that are whole number multiples of the test tone frequency. Values below 0.5% to 0.3% are negligible to the
untrained ear.

~1ExAs

1-18

INSTRUMENTS
POST OFFICE BOX 855303 • DALlAS, TEXAS 75265

AUDIO POWER AMPLIFIER
GLOSSARY
Power Supply Rejection Ratio (PSRR)
The log of the ratio of a change in supply voltage to the change in output power multiplied by 20. The result is
given in dB and measured at DC voltages. For example, the output of an audio power amplifier that has a PSRR
equal to 70 dB would change by 31.6 mV if the supply voltage changed by 0.1 V.
PSRR 20 x 10g(VsupplyNout) dB.

=

Crest Factor
The log of the ratio of peak output power to RMS output power multiplied by 10, typically given in decibels (dB).
This is commonly referred to as dynamic range. As the crest factor increases the difference between the peaks
and the normal loudness increases. Crest Factor = 10 x Log(PpEAWPRMS)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

1-19

1-20

2-1

Contents

o

ii"
In
In
I

C

l>

c

a.

Page
TPA2000D2 - 2-W Filterless Stereo Class-D Audio Power Amplifier .. 2-3
TPA005D12 - 2-W Class-D Stereo Audio Power Amplifier ............ 2-19
TPA005D14 - 2-W Class-D Stereo Audio Power Amplifier ............ 2-25
TPA005D02 - 2-W Class-D Stereo Audio Power Amplifier ............ 2-53
TPA032D01 - 10-W Class-D Mono Audio Power Amplifier ............ 2-77
TPA032D02 -10-W Class-D Mono Audio Power Amplifier ............ 2-97
TPA032D03 - 10-W Class-D Mono Audio Power Amplifier ........... 2-119
TPA032D04 -10-W Class-D Mono Audio Power Amplifier .......... 2-141

s·
"tJ

i...

l>

3

-CD·...._.

"'0

...

In

2-2

TPA2000D2
2-W FILTERLESS STEREO CLASS·D AUDIO POWER AMPLIFIER
• Modulation Scheme Optimized to Operate
Without a Filter
• 2 W Into 3-n Speakers (THD+N< 0.4%)
• < 0.08% THD+N at 1 W, 1 kHz, Into 4-n Load
• Extremely Efficient 3rd Generation 5-V
Class-D Technology:
- Low Supply Current (No Filter) •.. 8 mA
- Low Supply Current (Filter) ..• 15 mA
- Low Shutdown Current .•• 1 !lA
- Low Noise Floor ••• 5611VRMS
- Maximum Efficiency Into 3 n, 65 - 70%
- Maximum Efficiency into 8 n, 75 - 85%
- 4 Internal Gain Settings ••• 8 - 23.5 dB
- PSRR ... -77 dB
• Integrated Depop Circuitry
• Short-Circuit Protection (Short to Battery,
Ground, and Load)
• -40°C to 85°C Operating Temperature
Range

PWPPACKAGE
(TOP VIEW)
PGND
LOUTN
GAl NO
PVDD
LINN
AGND
eose
RINN
PVDD
SHUTDOWN
ROUTN
PGND

10
2
3
4
5
6
7
8
9
10
11
12

24
23
22
21
20
19
18
17
16
15
14
13

PGND
LOUTP
BYPASS
PVDD
L1NP
VDD
Rose
RINP
PVDD
GAIN1
ROUTP
PGND

description

The TPA2000D2 is the third generation 5-V class-D amplifier from Texas Instruments. Improvements to
previous generation devices include: lower supply current, lower noise floor, better efficiency, four different gain
settings, smaller packaging, and fewer external components. The most significant advancement with this device
is its modulation scheme that allows the amplifier to operate without the output filter. Eliminating the output filter
saves the user approximately 30% in system cost and 75% in PCB area.
The TPA2000D2 is a monolithic class-D power IC stereo audio amplifier, using the high switching speed of
power MOSFET transistors. These transistors reproduce the analog signal through high-frequency switching
of the output stage. The TPA2000D2 is configured as a bridge-tied load (BTL) amplifier capable of delivering
greater than 2 W of continuous average power into a 3-n load at less than 1% THD+N from a 5-V power supply
in the high fidelity range (20 Hz to 20 kHz). With 1 W being delivered to a 4-n load at 1 kHz, the typical THD+N
is less than 0.08%.
A BTL configuration eliminates the need for external coupling capacitors on the output. Low supply current of
8 mA makes the device ideal for battery-powered applications. Protection circuitry increases device reliability:
thermal, over-current, and under-voltage shutdown.
Efficient class-D modulation enables the TPA2000D2 to operate at full power into 3-n loads at an ambient
temperature of 85°C.
AVAILABLE OPTIONS
PACKAGED DEVICE
TA

TSSOP(PWP)

-40°C to 85°C

TPA2000D2PWP

NOTE: The PWP package is available taped and reeled. To
order a taped and reeled part, add the suffix R to the
part number (e.g., TPA2000D2PWPR).

•.

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments.
Copyright © 2000, Texas Instruments Incorporated

-!!1TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-3

TPA2000D2
2·W FILTERLESS STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS291 B - MARCH 2000 - REVISED APRIL 2000

functional block diagram
VDD

AGND

r--------------r---r-----------------,
VDD

RINN - - 1 - - - - _ + _...

Gate
Drive

PVDD

...-t-+-

ROUTN

'-+-+-

PGND

~--+-PVDD

RINP

Gate
Drive

--t----~H

.-t-+--

ROUTP

'--f---+- PGND

oc

SHUTDOWN
GAIN1
GAINO

Detect

nn

Biases
and
References

OC
Detect

Ramp
Generator"""

COSC~----+_-----+_~
ROSC~-----+-------+--~

BYPASS - - - - f - - - - _ + _ - - - - - - e
~--+-PVDD

LlNP - t - - - - - H H

Gate
Drive

.-t-+--

LOUTP

'-+-+-

PGND

~--+I- pVDD

LINN

Gate
Drive

--t----~H

I

I
...--+-- LOUTN
I
I

J
I~ ____________________________________
'-----+PGND

~TEXAS

INSTRUMENTS
2-4

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA2000D2
2-W FILTERLESS STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S291B - MARCH 2000 - REVISED APRIL 2000

Terminal Function
TERMINAL
NAME

I

NO.

110

DESCRIPTION

AGND

6

-

Analog ground

BYPASS

22

I

Tap to voltage divider for internal midsupply bias generator used for analog reference.

eose

7

I

A capacHor connected to this terminal sets the oscillation frequency in conjunction with ROSe. For proper
operation, connect a 220 pF capacitor from eose to ground.

GAINO

3

I

Bit 0 of gain control (TTL logic level)

GAINI

15

I

Bit 1 of gain control (TTL logic level)

LINN

5

I

Left channel negative differential audio input

L1NP

20

I

Left channel positive differential audio input

LOUTN

2

0

Left channel negative audio output

23

0

Left channel positive audio output

1,24

Power ground for left channel H-bridge

Right channel negative differential audio input

LOUTP

9, 16

-

RINN

8

I

RINP

17

I

Right channel positive differential audio input

I

A resistor connected to this terminal sets the oscillation frequency in conjunction with eose. For proper
operation, connect a 120 kn resistor from ROSe to ground.

PGND
PVDD

ROSe

12,13
4,21

18

Power ground for right channel H-brldge
Power supply for left channel H-bridge
Power supply for right channel H-bridge

ROUTN

11

0

Right channel negative audio output

ROUTP

14

0

Right channel positive output

SHUTDOWN

10

I

Places the amplifier in shutdown mode if a TTL logic low is placed on this terminal; normal operation if a TTL
logic high is placed on this terminal.

VDD

19

-

Analog power supply

absolute maximum ratings over operating free-air temperature (unless otherwise noted)t
Supply VOltage, Voo, PVoo ......................................................... -0.3 V to 6 V
Input voltage, VI ............................................................ -0.3 V to Voo+O.3 V
Continuous total power dissipation .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. See Dissipation Rating Table
Operating free-air temperature range, TA ............................................ -40°C to 85°C
Operating junction temperature range, TJ ........................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1116 inch) from case for 10 seconds ................................ 260°C

t Stresses beyond those listed under "absolute maximum ratings" may ceuse permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions· is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

TA S 25°C
POWER RATING

DERATING FACTOR
ABOVE TA 25°C

TA = 70°C
POWER RATING

TA = 125°C
POWER RATING

PWP

2.7W

21.8mWfOe

1.7W

1.4 W

=

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • OAUAS. TEXAS 75265

- - - - - _. .

~-.

2-5

TPA2000D2
2-W FILTERLESS STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S291 B - MARCH 2000 - REVISED APRIL 2000

recommended operating conditions
MIN
4.5

Supply voltage, VDD, PVDD

I GAl NO, GAIN1, SHUTDOWN
.1 GAl NO, GAIN1, SHUTDOWN

High-level input voltage, VIH
Low-level input voltage, VIL

MAX

UNIT

5.5

V

0.8

V

2

V

°c

Operating free-air temperature, TA

.,-40

85

PWM Frequency

200

300

kHz

TYP

MAX

UNIT

electrical characteristics, TA

=25°C, Voo =PVoo =5 V (unless otherwise noted)

PARAMETER

TEST CONDITIONS

IVool

Output offset voltage (measured differentially)

VI=OV

PSRR

Power supply rejection ratio

VDD=PVDD = 4.5 V to 5.5 V

IIH

High-level input current

VDD=PVDD = 5.5 V, VI = VDD = PVDD

IlL

Low-level input current

VDD=PVDD = 5.5Y, VI = 0 V

IDD

Supply current

No filter (with or without speaker load)

IDD

Supply current

With filter

IDDlsm

Supply current, shutdown mode

operating characteristics, TA
noted)

,L=22(.1H,

mV

1

(JA

10

mA

dB

-1

(JA
8

C= 1 (.IF

15
1

mA
10

(JA

=25°C, Voo =PVoo =5 V, RL =4 n, Gain =-2 VN (unless otherwise

PARAMETER

TEST CONDITIONS

MIN

Output power

THD=0.1%,

f=1 kHz,

THD+N

Total harmonic distortion plus noise

PO=1 W,

f= 20 Hz to 20 kHz

BOM

Maximum output power bandwidth

THD=5%

kSVR

Supply ripple rejection ratio

f= 1 kHz,

SNR

Signal-to-noise ratio

TYP

RL=3g

2

20
CIBYPASSI

=0.4 (.IF

20 Hz to 20 kHz, No input

Input impedance

Table 1_ Gain Settings
AMPLIFIER GAIN
(dB)

INPUT IMPEDANCE
(kQ)

TYP

TYP

8

104

1

12

74

1

0

17.5

44

1

1

23.5

24

GAINO

GAIN1

0

0

0

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

MAX

UNIT
W

<0.5%

--60
87

Integrated noise floor

2--6

10

-77

Po

ZI

MIN

kHz
dB
dBV

56

(.IV

>20

kg

TPA2000D2
2-W FILTERLESS STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S291B - MARCH 2000 - REVISED APRIL 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
11

THD+N

Efficiency

vs Output power

In-band output spectrum

vs Frequency

Total harmonic distortion plus noise

2,3

4

vs Output power

5-7

vs Frequency

8,9

test set-up for graphs
The THD+N measurements shown do not use an LC output filter, but use a low pass filter with a cut-off frequency
of 20 kHz so the switching frequency does not dominate the measurement. This is done to ensure that the
THD+N measured is just the audible THD+N. The THD+N measurements are shown at the highest gain for
worst case.
The LC output filter used in the efficiency curves (Figure 2 and 3) is shown in Figure 1.

= =

=

L1 L2 22 J.lH (DCR 110 mO,
Part Number = SCD0703T-220 M-S,
Manufacturer = GCI)
C1 = C2 = 1 J.lF
The ferrite filter used in the efficiency curves (Figure 2 and 3) is shown in Figure 1, where L is a ferrite bead.
L1
C1

=L2 =ferrite bead (part number =2512067007Y3, manufacturer =Fair-Rite)

=C2 =1 nF

OUT+

OUT-

Figure 1. Class-D Output Filter

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2-7

TPA2000D2
2-W FILTERLESS STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S291B - MARCH 2000 - REVISED APRIL 2000

TYPICAL CHARACTERISTICS
EFFICIENCY

EFFICIENCY

va

va

OUTPUT POWER

OUTPUT POWER

90

80

Ferrite Bead Filter
80

~

70

'#.
I

I

"

60

No Flit;'

f

50

20

/

~

/'

I(
o

r~

'#.

V

50

~~S-AB

r;

""
I

I

I

/

Notebook Speaker

c
.1

40

v---

m

30

./
V

u

RL = 8 n, Multimedia Speaker
VOO=5V
0.2

/

I

/ClaSS-AB

/'

20

10

o

L

60

..... ~
i-""""

40

30

LC Fitter

l/

,/

--

Ferrite Bead Filter

LC Filter

70

I

10

-

I

0.4
0.6
O.S
Po - Output Power - W

o

1.2

V

/

I

RL = 3 n, Notebook PC Speaker
VOO=5V
I

o

Figure 3

Figure 2
IN-BAND OUTPUT SPECTRUM
VDD=5V,

-2(111-+-+--+--+--+--+--1----11-- Gain = 8 dB,

fiN =fO = 1 kHz,

-40 1--t-+--+----1---+---+--+-_+_ Po =1.5 W,

m

"'i'

Bandwidth = 20 Hz to 22 kHz,
-60 1--t-+--+--.---1---+--=--+--+-_+_ 16386 Frequency Bins

i
o

2k

4k

6k

Sk

10k
12k
14k
f - Frequency - Hz

16k

Figure 4

2-a

:-II
TEXAS
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I

1.5
0.5
Po - Output Power - W

lSk

20k

22k 24k

2

TPA2000D2
2·W FILTERLESS STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS291 B - MARCH 2000 - REVISED APRIL 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER
10

10
VOD=~V

Gain = 23.5 dB
RL=3!l

'#.
I

c

I

~

i!

i

i

.2
c

~

-

:z:

~

0.1

I

z

c+
~

.~

1 kHz

.... .......

0

!

VOO=5V
Gain = 23.5 dB
r- RL=4!l

c

0

~
~

'#.

/

K .....,

.. ,

~

.....

:z:

fJ1

!

~

-

"

0.1

Z
+
C

'\.

:z:
I-

"

III

100m
Po - Output Power - W

20kHz

I

vs

OUTPUT POWER

FREQUENCY
10

r=

'#.

VOO=5V
Gain = 23.5 dB
RL=4!l

~

i!

s
is

i

.~

.2
c

1 kHz

0

!as

:t::t- .....

0.1

~'

'E
~

t

I

Z
+
C

20kHz
0.01
10m

Till

"

'-J

I"

0.75W

:z:

Z
+
C

:z:
I-

0.2W

!as

20Hz

:z:

~

3

=
=
-

I

c

0

!

2

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

VOO=5V
~ Gain = 23.5 dB
~ RL=S!l

c

........

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

'#.

r

'i: 17'=:::-

100m
Po - Output Power - W

Figure 5

10

"

J\

IIIIII

0.01
10m

2 3

2 Hz

/

I

20kHz
0.01
10m

1 kHz

0

20Hz

/

0.1

I....

..... ~

:z:
I-

\.

~

;7
0.01

100m
Po - Output Power - W

2

3

20

100

rt

~

1.5W

'oJ ~

~

i>'

II

I
1k

10 k 20 k

f - Frequency - Hz

Figure 8

Figure 7

-!!1 TEXAS

INSTRUMENTS
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2-9

TPA2000D2
2-W FILTER LESS STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS291 B - MARCH 2000 - REVISED APRIL 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY
10

~

VOO=5V
~ Gain = 23.5 dB
r- RL=sn

 OV

m m m m
I
I

I
I

Figure 11. The TPA2000D2 Output Voltage and Current Waveforms Into an Inductive Load
efficiency: why you must use a filter with the traditional class-D modulation scheme

The main reason that the traditional class-D amplifier needs an output filter is that the switching waveform
results in maximum current flow. This causes more loss in the load, which causes lower efficiency. The ripple
current is large for the traditional modulation scheme because the ripple current is proportional to voltage
multiplied by the time at that voltage. The differential voltage swing is 2 x Voo and the time at each voltage is
half the period for the traditional modulation scheme. An ideal LC filter is needed to store the ripple current from
each half cycle for the next half cycle, while any resistance causes power dissipation. The speaker is both
resistive and reactive, whereas an LC filter is almost purely reactive.
The TPA2000D2 modulation scheme has very little loss in the load without a filter because the pulses are very
short and the change in voltage is Voo instead of 2 x Voo. As the output power increases, the pulses widen
making the ripple current larger. Ripple current could be filtered with an LC filter for increased efficiency, but for
most applications the filter is not I)eeded.
An LC filter with a cut-off frequency less than the class-D switching frequency allows the switching current to
flow through the filter instead ofthe load. The filter has less resistance than the speaker that results in less power
diSSipated, which increases efficiency.

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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA2000D2
2-W FILTERLESS STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS291B - MARCH 2000 - REViseD APRIL 2000

APPLICATION INFORMATION
effects of applying a square wave Into a speaker
Audio specialists have said for years not to apply a square wave to speakers. If the amplitude of the waveform
is high enough and the frequency of the square wave is within the bandwidth of the speaker, the square wave
could cause the voice coil to jump out of the air gap and/or scar the voice coil. A 250-kHz switching frequency,
however, is not significant because the speaker cone movement is proportional to 1/12 for frequencies beyond
the audio band. Therefore, the amount of cone movement at the switching frequency is very small. However,
damage could occur to the speaker if the voice coil is not designed to handle the additional power. To size the
speaker for added power, the ripple current dissipated in the load needs to be calculated by subtracting the
theoretical supplied power, PSUPTHEORETICAL, from the actual supply power, PSUPo at maximum output power,
POUT. The switching power dissipated in the speaker is the inverse of the measured efficiency, 1'\MEASURED,
minus the theoretical efficiency, 1'\THEORETICAL.
PSPKR =PSUP - PSUP THEORETICAL (at max output power)

(1)

PSPKR = PSUP / POUT - PSUP THEORETICAL / POUT (at max output power)

(2)

PSPKR = lI1'\MEASURED - 1/1'\THEORETICAL (at max output power)

(3)

The maximum efficiency of the TPA2000D2 with an 8-n load is 85%. Using equation 3 with the efficiency at
maximum power from Figure 2 (78%), we see that there is an additional 106 mW dissipated in the speaker. The
added power dissipated in the speaker is not an issue as long as it is taken into account when choosing the
speaker.

when to use an output filter
Design the TPA2000D2 without the filter if the traces from amplifier to speaker are short. The TPA2000D2
passed FCC and CE radiated emissions with no shielding with speaker wires 8 inches long or less. Notebook
PCs and powered speakers where the speaker is in the same enclosure as the amplifier are good applications
for class-D without a filter.
A ferrite bead filter can often be used if the design is failing radiated emissions without a fiHer, and the frequency
sensitive circuit is greater than 1 MHz. This is good for circuits that just have to pass FCC and CE because FCC
and CE only test radiated emissions greater than 30 MHz. If choosing a ferrite bead, choose one with high
impedance at high frequencies, but very low impedance at low frequencies.
Use an output filter if there are low frequency « 1 MHz) EMI sensitive circuits and/or there are long leads from
amplifier to speaker.

gain setting via GAINO and GAIN1 Inputs
The gain of the TPA2000D2 is set by two input terminals, GAl NO and GAIN1.
The gains listed in Table 2 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, ZI, to be dependent on the gain setting. The actual gain settings are controlled
by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input
impedance may shift by 30% due to shifts in the actual resistance of the input resistors.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 20 kn, which is the absolute minimum input impedance of the TPA2000D2. At the higher gain
settings, the input impedance could increase as high as 115 kil.

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"

2-13

TPA2000D2
2·W FILTERLESS STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S291B - MARCH 2000 - REVISED APRIL 2000

APPLICATION INFORMATION
Table 2. Gain Settings
AMPLIFIER GAIN
(dB)

INPUT IMPEDANCE
(kn)

TYP

TYP

8

104

1

12

74

1

0

17.5

44

1

1

23.5

24

GAINO

GAIN1

0

0

0

input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. lfan additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r------------

I
I

Input

zF

~1---......--""_fI___'l,I\/Ir_~

Slgnal~

R

The -3 dB frequency can be calculated using equation 4:

f

-3 dB - 2n:

1

e,( R II ZI)

(4)

If the filter must be more accurate, the value of the capaCitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.

input capacitor, CI
In the typical application an input capacitor, el, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, el and the input impedance of the amplifier, ZI, form a
high-pass filter with the comer frequency determined in equation 5.

fC(hiQhpaSS) =

(5)

2lt~ICI

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TPA2000D2
2-W FILTERLESS STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS291B - MARCH 2000 - REVISED APRIL 2000

APPLICATION INFORMATION
The value of CI is important as it directly affects the bass (low frequency) performance of the circuit. Consider
the example where ZI is 20 kO and the specification calls for a flat bass response down to 80 Hz. Equation 5
is reconfigured as equation 6.

C I -

1

2:n:Z, fc

(6)

In this example, C, is 0.1 IlF so one would likely choose a value in the range of 0.1 IlF to 1 1lF. If the gain is known
and will be constant, use Z, from Table 1 to calculate C,. A further consideration for this capacitor is the leakage
path from the input source through the input network (C,) and the feedback network to the load. This leakage
current creates a dc offset voltage at the input to the amplifier that reduces useful headroom, especially in high
gain applications. For this reason a low-leakage tantalum or ceramic capacitor is the best choice. When
polarized capacitors are used, the positive side of the capacitor should face the amplifier input in most
applications as the dc level there is held at Vool2, which is likely higher than the source dc level. Note that it
is important to confirm the capacitor polarity in the application.
C, must be 10 times smaller than the bypass capacitor to reduce clicking and popping noise from power on/off
and entering and leaving shutdown. After sizing CI for a given cut-off frequency, size the bypass capacitor to
10 times that of the input capacitor.
C, s; CBYP / 10

(7)

power supply decoupllng, Cs
The TPA2000D2 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 IlF placed as close as possible to the device Voo lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 IlF or greater placed near
the audio power amplifier is recommended.

midrail bypass capacitor, CBYP
The midrail bypass capacitor, CBYP. is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CBYP. values of 0.471lF to 1 IlF ceramic or tantalum low-ESR capacitors are recommended
for the best THD and noise performance.
Increasing the bypass capaCitor reduces clicking and popping noise from power on/off and entering and leaving
shutdown. To have minimal pop, CBYP should be 10 times larger than C,.
CBYP ~ 10 x C,

(8)

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2-15

TPA2000D2
2-W FILTERLESS STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS291 B - MARCH 2000 - REVISED APRIL 2000

APPLICATION INFORMATION

differential input
The differential input stage of the amplifier cancels any noise that appears on both input lines of a channel. To
use the TPA2000D2 EVM with a differential source, connect the positive lead of the audio source to the RINP
(LlNP) input and the negative lead from the audio source to the RINN (LINN) input. To use the TPA2000D2 with
a single-ended source, ac ground the RINN and LINN inputs through a capacitor and apply the audio single to
the RINP and LlNP inputs. In a single-ended input application, the RINN and LINN inputs should be ac grounded
at the audio source instead of at the device inputs for best noise performance.

shutdown modes
The TPA2000D2 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, Ipp(SP) =1 ~. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.

using low-ESR capaCitors
Low-ESR capaCitors are recommended throughout this application section. A real (as opposed to ideal)
capaCitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capaCitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capaCitor.

evaluation circuit

r-rlF~~~r-----

GND

~

UN+

,-~r
'----l---=2'-1LOUTN
~t,-~~=--~----,!3~GAINO

Cl

1 PGND

o lC~

< 120 k

\1 C17

4 LPVDD

~i
Rll
II 0.1 ~F 5
LlN- /~t--\-+--'l---It------"-ILiNN
+---*------"6'-'1AGND

C3

RIN-

RIN+

~r

\1 C7

7 COSC

11220pF

8

/~

~

~.'~F

GND,
SHUTDOWN

>-

cIa \1
0.1 ~/I

~H
() f-=Sl

I

9 RPVDD
10 _ __
SHUTDOWN

~

r

'2

PGND

ROUTN
PGND

GND

4

LOUTP~~"-----+++---------

10

~F

1\

VDD

VDD

VDD

ROSC 18
17

RINN

R3
,,_____jL..2.20k!
~\=~

...c-<

-::!:--

TPA2000D2

1

"F

RINP 16

C20
0.1

~F

L.....

RPVDD 15 C19

I

R2 ..
120k

l:'o4k

C6 If
10uFI\

_ft. ' "

-=-

"

~

j1 ~

GAINI ""'4"-0-".,""?-::IFF'=+--+--+---~----'
ROUTP
PGND ",,'~"-----+--+

f

ROUT+
GND
GND
GND

L....----------------------
]J
0

:I:

(")

l;;
cp
C

J>

c:
C

8 5
0

"~

m

J>

PVcc
GENERATOR

I

I

I

~

G)

=e~
~8
mc.n

:2J

VDDf-VDD

COSC

iii
~

I
I
I

10ka

I

I~g
"3:
~"'

•

DD

LPVDD
,•

LCOMP I

!!1

0-4r
~ ~ ~.

VCPPV

i3 i

~

c

1\)-1

~

:l!

c CI
z!CI

(50000

'V:~~]

NOTE A. LPVDD. RPVDD. VDD. and PVDD are externally connected. AGND and PGND are externally connected.

....==

"

:;;
in
:2J

TPA005D12
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S241 B - AUGUST 1999 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME

DESCRIPTION

NO.

AGNO

3,7,20,
46,47

COSC

48

Capacitor I/O for ramp generator. Adjust the capacitor size to change the switching frequency.

CP1

25

First diode node for charge pump

CP2

24

First inverter switching node for charge pump

CP3

23

Second diode node for charge pump

CP4

26

Second inverter switching node for charge pump

FAULTO

42

Logic level faultO output signal. Lower order bit of the two !au~ signals w~h open drain output.

FAULT1

41

Logic level fau~1 output signal. Higher order bit of the two !auM signals with open drain output.

LCOMP

6

Compensation capacitor terminal for left·channel Class·O amplifier

LINN

4

Class-O left-channel negative input

LINP

5

Class-O left-channel positive input

Analog ground for headphone and Class-O analog sections

LOUTN

14,15

Class-O amplifier left-channel negative output of H-bridge

LOUTP

10,11

Class-O amplifier left-channel positive output of H-bridge

LPVOO

9,16

Class-O amplifier left-channel power supply

MUTE

2

NC

17,18,19,
30,31,32

Active-low logic-level mute input signal. When MUTE is held low, the selected amplifier is muted. When MUTE
is held high, the device operates normally. When the Class-D amplifier is muted, the low-side output transistors
are turned on, shorting the load to ground.
No connection
Power ground for left-channel H-bridge only

PGNO

12,13

PGNO

27

PGNO

36,37

Power ground for right-channel H-bridge only

PVOO

21,28

VDD supply for charge-pump and gate-drive Circuitry

Power ground for charge pump only

RCOMP

43

Compensation capacitor terminal for right-channel Class-O amplifier

RINN

45

Class-D right-channel negative input

RINP

44

Class-O right-channel positive input

RPVOO

33,40

Class-O amplifier right-channel power supply

ROUTN

34,35

Class-O amplifier right-channel negative output of H-bridge

ROUTP

38,39

Class-O amplifier right-channel positive output of H-bridge

SHUTOOWN

1

Active-low logic-level shutdown input signal. When SHUTDOWN is held low, the device goes into shutdown mode.
When SHUTOOWN is held at logic high, the device operates normally.

V2P5

29

2.5-V intemal reference bypass

VCP

22

Storage capacitor terminal for charge pump

VOD

8

VOO bias supply for analog circuitry. This terminal needs to be well filtered to prevent degrading the device
performance.

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TPAOO5D12
2-W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS241 B - AUGUST 1999 - REVISED MARCH 2000

Class·D amplifier faults
Table 1. Class-O Amplifier Fault Table
FAULTot

FAULT1t

1

1

No fault. - The device is operating normally.

1

Charge pump under-voltage lock-out (VCP-UV) fault. - All low-side transistors are tumed on, shorting the load to
ground. Once the charge pump vo~age is restored, normal operation resumes, but FAULT1 is still active. FAULT1 is
cleared by cycling MUTE, SHUTDOWN, or the power supply.

1

0

Over-current fault. - The output transistors are all switched off. This causes the load to be in a high-impedance state.
This is a .Iatched fault and is cleared by cycling MUTE, SHUTDOWN, or the power supply.

0

0

Thermal fault. - All the low-side transistors are tumed on, shorting the load to ground. This is latched fault and is
cleared by cycling MUTE, SHUTDOWN, or the power supply.

0

-

DESCRIPTION

t These logiC levels assume a pullup to PVDD from the open-drain outputs.

=

absolute maximum ratings over operating free-air temperature range, TC 25°C (unless otherwise
noted)*
Supply voltage, Vpp (PVpp, LP~PVpp, Vpp) ........................................... 5.5 V
Input voltage, VI (SHUTDOWN, MUTE) ............................................. -0.3 V to 5.8 V
Output current, 10 (FAULTO, FAULT1), open drain terminated ................................... 1 mA
Charge pump voltage, Vcp .......................................................... PVpp + 15 V
Continuous H-bridge outP!Jt current .......................................................... 2 A
Pulsed H-Bridge output current, each output, Imax (see Note 1) .................................. 5 A
Continuous total power dissipation .................................... See Dissipation Ratings Table
Operating virtual junction temperature range, TJ .................................... -40°C to 150°C
Operating case temperature range, T C ............................................ -40°C to 125°C
Storage temperature range, Tstg .................................................. -40°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: Pulse duration = 10 ms, duty cycle :5 2%
DISSIPATION RATING TABLE

t

=

=

=

PACKAGE

TAS25°Ct
POWER RATING

DERATING FACTOR
ABOVE TA 25°C

TA 70°C
POWER RATING

TA 85°C
POWER RATING

TA 125°C
POWER RATING

DCA

5.6W

44.8mW/oC

3.6W

2.9W

1.1 mW

=

Pleese see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (literature number
SLMA002), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the
information in the section entitled Texas Instruments Recommended Board for PowerPADon page 33 of the before mentioned
document.

recommended operating conditions
MIN
4.5

Supply voltage, PVDD, LPVDD, RPVDD, VDD
High-level input voltage, VIH

MAX
5.5

0.75
1

Audio inputs, LINN, LlNP, RINN, RINP, differential input voltage
PWM frequency

150

~lExA.s

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

UNIT
V
V

4.25

LOW-level input voltage, VIL

2-22

NOM

450

V
VRMS
kHZ

TPA005D12
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS241 B - AUGUST 1999 - REVISED MARCH 2000

=

=

electrical characteristics, Class-D amplifier, VDD PVDD LPVDD
TC = 25°C, See Figure 1 (resistive load) (unless otherwise noted)
PARAMETER

=RPVDD =5 V, RL =4 n,
MIN

TEST CONomONS
VOO = PVOO = LPVOO = RPVOO = 4.5 V to
5.5V

TVP

MAX

UNIT

PSRR

Power supply rejection ratio

100

Supply current

No load,

25

35

mA

IOO(MUTE)

Supply current, mute mode

MUTE=OV

3.9

10

mA

IOO(SO)

Supply current, shutdown mode

SHUTDOWN = 0 V

0.2

10

IIH

High-level input current

VIH =5.3V

IlL

Low-level input current

VIL=-0.3V

-1

I1A
I1A
I1A

roS(on)

Total static drain-to-source on-state
resistance (low-side plus high-side FETs)

IO=2A

900

mO

roS(on)

Matching, high-side to high-side, low-side to
low-side, same channel

IO=0.5A

operating characteristics, Class-D amplifier, VDD
TC = 25°C, See Figure 1 (unless otherwise noted)

Noliltar

1

700
95%

99%

=PVDD =LPVDD =RPVDD =5 V, RL =4 n,

PARAMETER

TEST CONomONS

MIN

TYP

Po

RMS output power, THO = 0.5%, per channel

THO+N

Total hannonic distortion plus noise

PO=l W,

1=1 kHz

0.2%

Efficiency

PO=l W,

RL=BO

BO%

AV

Gain

MAX

25
95%

dB

99%

-55

Noiseffoor

dBV

70

Dynamic range
Crosstalk
Frequency response bandwidth, post output fiHer, -3 dB

dB

-55

f= 1 kHz

UNIT
W

2

Left/right channel gain matching

BOM

dB

40

20

Maximum output power bandwidth

dB
20000

Hz

20

kHz

thermal resistance
PARAMETER
RSJP

TEST CONOmONS

MIN

TVP

Thennal resistance, junction-to-pad
Thennal shutdown temperature

165

MAX

UNIT

10

°CIW
°C

~TEXAS

INSTRUMENTS
POST OFF1CE BOX 655303 • DAUAS, TEXAS 75265

2-23

TPA005D12
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S241 B - AUGUST 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r-------------------,
I

FAULTO~

I

FAULT1

I
~
2 I
---:1
~
-=I

pVDD

SHUTDOWN
MUTE

PVDD

5 V 9,16
11lF
Balanced
Differential
Input Signal

I

I

tll-

I
151lH
LOUTN;.-I1.:..:4,,-,,1.::..5-fYYY"'......_ _......._ - - - ,

AGND
LPVDD

I
I

{-j~

LlNP

~~
I~ LINN

11lF

n

6 I LCOMP
43 I
RCOMP

I
470PF--L

J

470 PF T

1

r-1
-=-

470 PF T

11lF

-

r::..:._t-l

I

_~~

=-=_-,t

I

1 RPVDD
AGND (see Note A)
1 PGND (see Note A)

l

T

47 nF

2,21lF

I
I

5V~PVDD

I,
I
I
I
I
I

ROUTP 1-38
=39!!........1YY''"'--4t--_

I

I
I
~-------------------~

Figure 1, 5-V, 4-Q Test Circuit, Class-O Amplifier

~TEXAS

2-24

47nF

RINP

1 \ 1 RINN

33,40
7,20,46,47
12,13,27,36,37

±

!

{-j~
11lF

5V

cosc

I

-=Balanced
Differential
Input Signal

I

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

_*----'

TPAOOSD14
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
DCA PACKAGE
(TOP VIEW)

• Choose TPA2000D2 For Upgrade
• Extremely Efficient Class-D Stereo
Operation
• Drives Land R Channels, Plus Stereo
Headphones
•
•
•
•
•
•
•

2-W BTL Output Into 4 0
S-W Peak Music Power
Fully Specified for S-Y Operation
Low Quiescent Current
Shutdown Control ••• 0.2 IJA
Class-AB Headphone Amplifier
Thermally-Enhanced PowerPADTM Surface
Mount Packaging
• Thermal, Over-Current, and Under-Yoltage
Protection

description

SHUTDOWN
MUTE
MODE
LINN
LlNP
LCOMP
AGND
Voo
LPVoo
LOUTP
LOUTP
PGND
PGND
LOUTN
LOUTN
LPVoo
HPDL
HPLOUT
HPLIN
AGND
PVoo
VCP
CP3
CP2

10

48

2

47

3
4

46
45
44

5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21

43
42
41
40
39
38

37
36
35
34

33
32

31
30

COSC
AGND
AGND
RINN
RINP
RCOMP
FAULTO
FAULT1
RPVoo
ROUTP
ROUTP
PGND
PGND
ROUTN
ROUTN
RPVoo
HPDR
HPROUT
HPRIN
V2P5
PVoo
PGND
CP4
CP1

The TPA005D14 is a monolithic power Ie stereo
29
audio amplifier that operates in extremely efficient
28
Class-D operation, using the high switching speed
22
27
of power DMOS transistors to replicate the analog
23
26
input signal through high-frequency switching of
24
25
the output stage. This allows the TPA005D14 to
be configured as a bridge-tied load (BTL) amplifier
capable of delivering up to 2 W of continuous
average power into a 4-0 load at 0.4% THD+N from a 5-V power supply in the high-fidelity audio frequency
range (20 Hz to 20 kHz). A BTL configuration eliminates the need for external coupling capacitors on the output.
Included is a Class-AB headphone amplifier with interface logic to select between the two modes of operation.
Only one amplifier is active at any given time, and the other is in power-saving sleep mode. Also, a chip-level
shutdown control is provided to limit total quiescent current to 0.2 ~, making the device ideal for
battery-powered applications.
A full range of protection circuitry is included to increase device reliability: thermal, over-current, and
under-voltage shutdown, with two status feedback terminals for use when any error condition is encountered.
The high switching frequency of the TPA005D14 allows the output filter to consist of three small capacitors and
two small inductors per channel. The high switching frequency also allows for good THD+N performance.
The TPA005D14 is offered in the thermally enhanced 48-pin PowerPAD TSSOP surface-mount package
(designator DCA).

A

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

2-25

-r

i')

~

()

C»

I""

c

~ ~

LPVDD

LPVDD

I
I
I

UNPI

i~~

~~

LPVDD
:-

10kn

CONTROL and
LOGIC

•

-=-

I

RINPI

I

I

I MUTE
I

~
:z:

~

~

):.

e•
c:
S!

):.

0

"til

0

::E
m

GENERATOR

B:

-=-

:;;
iii

"til

VCP

PVDD

:::D

RPVDD

I HPROUT

I
IIHPRIN

I
RPVDDILRPVDD

Ir---:!:.
-

(I)
(I)

c

VCP-UVLO
DETECT

GATE
DRIVE

I

RINNI

AGND

~
en

roo

I

rT- 1.5V
~ ~ 10 kn
•

:::D .....
~

i

m

OVER·I
.---~
.....I DETECT

RAMP
GENERATOR

110 kn

-t~

me

n
roo

MODE

I
I

=eJ

(1)0

):.

n

I

I

c:
>
G)
c:

N-t

:::D

I
I

RCOMPI

I

()

I

f t u • . - _... _ .

GATE
DRIVE

VDDr-- VDD

COSC

I

II)

!!l

I
STARTUP

I
I

I

I
I
I

THERMAL
DETECT

GATE
DRIVE

1.5V

LCOMP(

~-~

~~~
)( :ad

UD

•

110kn

g

a 5 ----,

r-----------------------------------------i
I

f ~

il! cil!

6

0

f:

IL.. _ _ _PGND
________ ____

GATE
DRIVE

__________~

_

~
HP
DEPOP

I

HPDL

g _______ .J HPDR

c

~

~
~
c C

ZlC

Ci () () ()

-ol!;!;S

NOTE A. LPVOO. RPVOO. VOO. and PVOO are externally connected. AGNO and PGNO are externally connected.

...

..

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME

DESCRIPTION

NO.

AGND

7,20,
46,47

COSC

48

CapaCitor I/O for ramp generator. Adjust the capacitor size to change the switching frequency.

CPl

25

First diode node for charge pump

CP2

24

First inverter switching node for charge pump

CP3

23

Second diode node for charge pump

Analog ground for headphone and Class-D analog sections

CP4

26

Second inverter switching node for charge pump

FAULTO

42

Logic level lauHO output signal. Lower order bit of the two fauH signals with open drain output.

FAULTl

41

Logic level fauHl output signal. Higher order bit of the two fault signals with open drain output.

HPDL

17

Depop control for left headphone

HPDR

32

Depop control for right headphone

HPLIN

19

Headphone amplifier left input

HPLOUT

18

Headphone amplifier left output

HPRIN

30

Headphone amplifier right input

HPROUT

31

Headphone amplifier right output

LCOMP

6

Compensation capacitor terminal for left-channel Class-D amplifier

LINN

4

Class·D left-channel negative input

L1NP

5

Class-D left-channel positive Input

LOUTN

14,15

Class-D amplifier left-channel negative output of H-bridge

LOUTP

10,11

Class-D amplifier left-channel positive output of H-bridge

LPVDD

9,16

Class-D amplifier left·channel power supply

MODE

3

Logic-level mode input signal. When MODE is held low, the main Class-D amplifier is active. When MODE is held
high, the head phone amplifier is active.

MUTE

2

Active-low logic-level mute input signal. When MUTE is held low, the selected amplifier is muted. When MUTE is
held high, the device operates normally. When the Class·D amplifier is muted, the low-side output transistors are
turned on, shorting the load to ground.

PGND

12,13

PGND

27

PGND

36,37

Power ground for right·channel H-bridge only

PVDD
RCOMP

21,28

VDD supply for Charge-pump and gate-drive circuitry

Power ground for left-channel H-bridge only
Power ground for charge pump only

43

Compensation capacitor terminal for right-channel Class-D amplifier

RINN

45

Class-D right-channel negative input

RINP

44

Class-D right-channel positive input

RPVDD

33,40

Class-D amplifier right-channel power supply

ROUTN

34,35

Class·D amplifier right-channel negative output of H-bridge

ROUTP

38,39

SHUTDOWN

1

Class-D amplifier right-channel positive output of H-bridge
Active-lOW logic-level shutdown input signal. When SHUTDOWN is held low, the device goes into shutdown mode.
When SHUTDOWN is held at logic high, the device operates normally.

V2P5

29

VCP

22

Storage capacitor terminal for charge pump

8

VDD bias supply for analog circuitry. This terminal needs to be well filtered to prevent degrading the device
performance.

VDD

2.5-V Internal reference bypass

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-27

TPAOO5D14
2-W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

Class·D amplifier faults
Table 1. Class-D Amplifier Fault Table
FAULTot

FAULT1t

1

1

No fault. -

DESCRIPTION

0

1

Charge pump under-voltage lock-out (VCP-UV) fault - All low-side transistors are turned on, shorting the load to
ground. Once the charge pump voltage is restored, normal operation resumes, but FAULT1 is still active. FAULT1 is
cleared by cycling MUTE, SHUTDOWN, or the power supply.

1

0

Over-current fault - The output transistors are all switched off. This causes the load to be in a high-impedance state.
This is a latched fault and is cleared by CYCling MUTE, SHUTDOWN, or the power supply.

0

0

Thermal fauH - All the low-side transistors are turned on, shorting the load to ground. This is latched fault and is
cleared by cycling MUTE, SHUTDOWN, or the power supply.

The device is operating normally.

tThese logic levels assume a pullup to PVDD from the open-drain outputs.

headphone amplifier faults
The thermal fault remains active when the device is in head phone mode. This fault operates exactly the same
as it does for the Class-O amplifier (see Table 1).
If LPVoo or RPVoo drops below 4.5 V, the headphone is disabled by the under-voltage lockout circuitry. Once
LPVoo and RPVoo exceed 4.5 V, the headphone amplifier is re-enabled. No fault is reported to the user.
AVAILABLE OPTIONS
PACKAGED DEVICES
TA

TSSOpt
(DCA)

-40°C to 125°C

TPAOO5D14DCA

t The DCA package is available in left-ended tape and reel. To order
a taped and reeled part, add the suffix R to the part number (e.g.,
TPAOO5D14DCAR).

~TEXAS

2-28

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range, TC = 25°(; (unless otherwise
noted)*
Supply voltage, Voo (PVoo, LPVoo, RPVoo, Voo) ........................................... 5.5 V
Input voltage, VI (SHUTDOWN, MUTE, MODE) ...................................... -0.3 V to 5.8 V
Output current, 10 (FAULTO, FAULT1), open drain terminated ................................... 1 mA
Charge pump voltage, Vcp .......................................................... PVoo + 15 V
Continuous H-bridge output current .......................................................... 2 A
Pulsed H-Bridge output current, each output, Imax (see Note 1) .................................. 5 A
Continuous total power dissipation .................................... See Dissipation Ratings Table
Operating virtual junction temperature range, TJ .................................... -40°C to 150°C
Operating case temperature range, T C ............................................ -40°C to 125°C
Storage temperature range, Tstg .................................................. -40°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: Pulse duration = 10 ms, duty cycle s 2%
DISSIPATION RATING TABLE
PACKAGE

TA s 25°ct
POWER RATING

DCA

5.6W

=

DERATING FACTOR
ABOVE TA 25°C

=

TA 70°C
POWER RATING

TA = 85°C
POWER RATING

TA = 125°C
POWER RATING

3.6W

2.9W

1.1 mW

:I: See

the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (literature number
SLMA002), for more information on the PowerPAO package. The thermal data was measured on a PCB layout based on the
information in the section entitled Texas Instruments Recommended Board for PowerPAD on page 33 of the before mentioned
document.

recommended operating conditions
MIN

NOM

4.5

Supply voltage, PVOO, LPVOO, RPVOO, VOO
High-level input voltage, VIH (MUTE. MODE, SHUTDOWN)

5.5

0.75

Audio inputs, LINN, L1NP, RINN, RINP, HPLlN, HPRIN, differential input voltage

1

PWM frequency

150

=

electrical characteristics, Class-D amplifier, VDD PVDD
TA = 25°C, See Figure 1 (unless otherwise noted)

UNIT
V
V

4.25

Low-level input voltage, VIL (MUTE, MODE, SHUTDOWN)

PARAMETER

MAX

450

V
VRMS
kHZ

=LPVDD =RPVDD =5 V, RL =4 n,

TEST CONDITIONS

MIN

TYP

MAX

UNIT

Power supply rejection ratio

VOO = PVOO = LPVOO = RPVOO = 4.5 V to 5.5 V

100

Supply current

No output filter connected

25

35

rnA

IOO(MUTE)

Supply current, mute mode

MUTE=OV

3.9

10

rnA

IOO(SO)

Supply current, shutdown mode

SHUTDOWN = 0 V

0.2

10

IIH

High-level input current

VIH = 5.3 V

IlL

Low-level input current

VIL=-0.3V

-1

!1A
!1A
!1A

rOS(on)

Total static draln-to-source on-state
resistance (low-side plus high-side
FETs)

IO=0.5A

900

mQ

rOS(on)

Matching, high-side to high-side,
low-side to low-side, same channel

IO=0.5A

-40

dB

1

700

95%

98%

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

2-29

TPA005D14
2-W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

operating characteristics, Class-D amplifier, Voo = PVoo = LPVoo = RPVoo = 5 V, RL = 4 0.,
TA 25°C, See Figure 1 (unless otherwise noted)

=

PARAMETER

TEST CONDITIONS

MIN

TYP

Po

RMS output power

1= 1 kHz,
Per channel

THO = 0.5%,

THO+N

Total harmonic distortion plus noise

PO=1 W,

1=1 kHz

0.2%

Efficiency

PO=1 W,

RL=80

80%

Gain

AV

MAX

2

W

dB

20

Left/right channel gain matching

95%

Noisefioor

99%
dBV

-55
70

Dynamic range
Crosstalk

1= 1 kHz

BaM

Maximum output power bandwidth

ZI

Input impedance

dB

-55

Frequency response bandwidth, post output filter, -3 dB

UNIT

20

dB
20000

Hz

20

kHz

10

kQ

electrical characteristics, headphone amplifier, PVoo = LPVOO= RPVoo = 5 V, RL = 32 0., TA = 25°C,
See Figure 3 (unless otherwise noted)
PARAMETER

TEST CONDITIONS

Power supply rejection ratio

MIN

PVOO = 4.5 V to 5.5 V,
AV=-1 VN

Uncompensated gain range

TYP

MAX

-10

VN

8

10

mA

Supply current, mute mode

1.5

2

rnA

IOO(SO)

Supply current, shutdown mode

0.2

10

liB

Input bias current

30

IIA
IIA

100

Supply current

IOO(MUTE)

-1

UNIT
dB

-60

operating characteristics, headphone amplifier, PVoo = LPVoo = RPVoo = 5 V, RL = 32 0., TA = 25°C,

See Figure 3 (unless otherwise noted)
PARAMETER
Po

TEST CONDITIONS

Output power

THO =0.5%,
AV=-10VN

Supply voltage rejection ratio

1=1 kHz

MIN

1=1 kHz,

Noise floor
Dynamic range
Crosstalk

1=1kHz

Frequency response bandwidth, post output filter, -3 dB
BOM

Maximum output power bandwidth

ZI

Input impedance

TYP

MAX

50

mW

-60
-84

dBV

90

dB

dB

dB

-38
20

20000

Hz

20

kHz
MO

>1

thermal shutdown
PARAMETER

TEST CONDITIONS

Thermal shutdown temperature

MIN

TYP

165

~1EXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

UNIT

MAX

TPA005D14
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r-------------------,

1

FAULTO~

1

FAULT1~

1

1

--.!J SHUTDOWN
2 1 MUTE
-:-1
~ MODE
-=
1

PVDD

1
1511H
15
LOUTN:--1.:....:4.<..:..
=--.J-y""'''-4....-_---e-_----,
'

PVDD

Balanced
Dlfferantlal
Input Signal

5 V 9•16

I

ll1F

1
1

{----1~,

- - 1~

LPVDD

LlNP

LINN
ll1F
.--_ _----"6'-!1 LCOMP

470

PF~T

43 1 RCOMP

!
r4
!

470 PF ±

1

-=

casc

470 PFT

1

-=
ll1F

Balanced
Dlfferantlal
Input Signal

1

r-::::-'---'+ 47nF

1
{----1~, RINP
----1~ RINN
ll1F 1

;-=_---'+47nF

I

5V

33,40 RPVDD
7,20,46,47 1 AGND (see Nota A)
112,13.27,36.37 PGND (see Note A)

J-

lT

i

1
1

O.lI1F

5V~PVDD

~ HPLIN
1

i: I
30

-

HPRIN

1
1

ROUTP 1-'38=39~-y,,-,"-4....-_---e-_--'

1

1
1
~-------------------~

Figure 1. 5-V, 4-0 Test Circuit, Class-O Amplifier

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-31

TPA005D14
2·W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r---------------,

--J..,

5V

r:-

SHUTDOWN

~

FAULTO I 42

5V
MUTE
5 V _3_1 MODE

FAULT1 1

I
5 V~ LPVDD
5 I
I LlNP

.1

1 14,15
LOUTN ~

I

LOUTP j....!!!J.L

~ LINN

n

-=- 6 I
1'-----"--11
470 pF --LT

1

LCOMP

II

RCOMP

m~1 1
I

- 48

--L
470 pF T

-

-

I cosc

VDDr- 5V
I
HPLOUT 1-11-'-"S'---_22_0-'-I1_F

I

I

124
_
CP2 ;-:1"-.:-----1

117
HPDLi-- ' - ' - - - - - - - - - - - '
1

I
I
I

123
CP3 1-'1
I
CP41-'12=6_-,_

100 kO I

Vcp;-:I2=2_--,

I

Left SE
HP Input

.--J

RlghtSE
HPlnput

---1~0

19
100kn
100kn

0.111f 100
kO
HPROUT

=-----'l

T

I

HPLIN

I
I
I

I
I
HPRIN

I
1
ROUTN I 34,35

I
1

I
ROUTP I 3S,39
IL _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ JI

Figure 2. Headphone Test Circuit

~TEXAS

INSTRUMENTS
2--32

...iT 47 nF -=-

I

7,20,46,47 I AGND
12,13,27,36,37 1 PGND

HPLOUT

320
32 0

CP11-'12=5_--,

5V~PVDD

~

I~

I

~ RINN

220l1F

---ll

HPROUTII:~
HPDR

I
33,40 I RPVDD

5V

J1~

Is

44 I
I RINP

-=-

±

1

I
II

.1-=-

29

V2P5n

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

I

47 nF

--L
TO.1I1F

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT iNFORiviATiON
5V
To System {4;H-;;~~;----------'
Control
2
100 kQ
100kQ
100 kg
~ MUTE
1 42
+----'\N\.
3 I MODE
FAULTOi---'==-----+---e--}
1
1 41
To System
5V
::t:!:
:!:9.16 1 LPVDD
FAULT1
Control
10 J1F
-::r;- 1 I1F -::r;- 1 I1F 1
1

~ V

V I I
111F

Left Class-D Balanced
Differential Input
Signal

LOUTN 1---'-14~.1:c:.5_r~y"'_4t__-----<._-_,

1

{ -1~
-1~

I

LlNP

!

LINN

1

111F
. -_ _ _-"6'-11 LCOMP
470PF*
..L

-=-

43 1
- L I RCOMP
P
470 F r 1
-=- 1
r----=48=-!1 COSC

470PFrL

HPROUT 1-13".,1'----_ _-'--1
HPDR I32

17'--_ _ _ _t -..
HPDL r-:
1

RINN

1J1F
5V

1

::t:!:
£3.40
10I1F---r ~1I1F~1I1F

\J

7 20 46 47

RPVDD

r---------''"'''''''=~

1213273637

5V:!:

21.28

100kQ

1

{ -1~ RINP
-1~

5V

HPLOUT~ f--===-.t=----e---~

!
1

1 I1F
Right Class-D Balanced
Differential Input
Signal

1
1511H
V2P51-12",9=-------l'
18
5vT 111F
VDDI
:!:
1 1 I1F-::;r

1

-=-

4Q

!
LOUTP t--'-'10"-'1'-!.1_f"YY"I"__4a--____<._--'

CP1 !-12=5=------,
1

AGND

CP2 r-h=4,-----,
1

PGND
PVDD

CP31-12",3=------,
1
-l..-T 47nF

1 I1F ~ HPLOUT

CP41--!2=6=--------'

VCP~I~22~--------_,

l

1

LeftSE
HPlnput

--11---'\f\/\r-+--'-"--! HPLIN

RlghtSE
HPlnput

--11---'\f\/\r-+------'''''--! HPRIN

"I

1

0.1I1F

ROUTN~34~3~5-f"YY"I"--4t__-----<__--,~

1

!
1

O.

22I1F

h

0.2211F

4Q
-

ROUTP~38~3~9-f"YY)"__4a------<~--'

HPROUT

L _______________ ~

1

NOTE A.

~ = power ground and

-b

= analog ground

Figure 3. TPA032D04 Typical Configuration Application Circuit

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-33

TPA005D14
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Switching frequency

100

Supply current

THO+N

Total harmonic distortion plus noise

4
5,6

vs Fre&-air temperature

7,9,11
12,14,15
8,10,13

vs Frequency
vs Output power

Gain and phase

vs Frequency

16,17

Crosstalk

vs Frequency

18

Power dissipation

vs Output power

19

Efficiency

vs Output power

20

SUPPLY CURRENT

SUPPLY CURRENT

va

va

SWITCHING FREQUENCY

FREE-AIR TEMPERATURE

50

50

Class-D AmplHlar

I

Class-D AmPllfijr

~

40

I

'E

i

",

"

'"

WIth Output FiRer -

E

>-

is.
Q.

i

:s

(J

30

~

~~

:s

I/)

C

20

,

30

8:

:s

".

--

V

~

~

,......,.

---V

I/)

I

,SI

With Output Filter

40

I

(J

J

I

. / Without Output Filter

C

,SI

20
Without Output Filter

10
100

200

300

400

500

10
-50

f - Frequency - kHz

-25

0

I

I

I

50

75

100

TA - Free-Air Tempereture - °C

Figure 4

FigureS

~1ExAs

2-34

25

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265.

125

150

TPA005D14
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY CURRENT

TOTAL HARMONIC DISTORTION + NOISE

vs

vs

FREE-AIR TEMPERATURE

FREQUENCY

10.0
Class·D Amplifier
VDD=5V
RL=8Q

"/I.

Headphone Amplifier

I

9.5
9.0

!z

I

8.5

c

C
~

8.0

1i
0

c(

E

~

--

1W

II

......

/

500mW

0

'f

..

7.5

0.1

'c0

a.
a.
~
rn 7.0
I

0 6.5
_0

i

100mW

:z:

",.

~

1.1' ....

Y

0

~

0

-- --

+

-' ~

S

..",

6.0

~

5.5

Z
+
0

I

:z:

I-

5

0.01

-50

-25

0

25

50

75

100

125

150

20

100

1k

TA - Free-Air Temperature - °C

Figure 6

2

TOTAL HARMONIC DISTORTION + NOISE

vs

vs

OUTPUT POWER

FREQUENCY

Cla_D Amplifier
VDD=5V
RL=8Q

I

J!0

"/I.
I

+

i
0

0.1

~ r"l'

is
.S!
c

f= 20 kHz

~

f=20Hz

Lw.

I

i!=

I

0.1

10

~
11

,

~

IIIII
Class·D Amplifier
VDD=5V
RL=4Q

~

I"

0.02
0.01

-

l...I'

~

.......

~

~ ~mw

V

0.1 t'"

Is

LI7,;~ ~Hz
.J... .....

Z
+
0

V

i

II

'ii

~I

i!=

1W

+

5

~

.,E

./

z

c

:z:

2W

=

'0

z

.S!
c

30k

Figure 7

TOTAL HARMONIC DISTORTION + NOISE

"/I.

10k

f - Frequency - Hz

0.01
20

Po - Output Power - W

100

1k

10k

30k

f - Frequency - Hz

Figure 8

Figure 9

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-35

TPAOO5D14
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION + NOISE

TOTAL HARMONIC DISTORTION + NOISE

vs

vs

OUTPUT POWER

FREQUENCY

2

Class.. O Amplifier
VOO=5V
RL=4Q

'#.
I
CD

~
Z

f=

'#.
I

.~

-

Z

+

+
c

c

~0

~

~

ic

..
~

~
::t:

~

S

~I

0.1

_

~~

.JIll

~ ~=11 ~nr

~
AV=1

Ll..

c

f=20kHz

CO= 47O ILF

0.1

.2

'c0

Headphone Amplifier

t- CI = 10 ILF
t- RL=32Q

1/

0

..
~

r-I

::t:

S

~

~

0.01

/

Av=

I\.

\ \
~

1; =1

I

z
c+
j!:

r--- t- f=20Hz
r--1-

0.04
0.01

~

Z

,..,

I~

+

C

::t:
....

I I 111111
10

0.1

0.006
20

f - Frequency - Hz

Figure 10

Figure 11

TOTAL HARMONIC DISTORTION + NOISE

I

j
z0

+

c

~

~..

vs

FREQUENCY

OUTPUT POWER

I

.~

Z

+

c

~0

~

'c
0

~I

z+

C

::t:
....

-

'"

S

0.01
0.005
20

100

Headphone Amplifier
VOO=5V
AV=1
CI = 10 ILF
RI= RF=10kQ
Co = 470 ILF
f=20kHz

'#.

0.1

~
::t:

TOTAL HARMONIC DISTORTION + NOISE

vs
Headphone Amplifier
VO=1 V
PO=40mW
AV=1
CI = 10 ILF
RI= RF= 10kQ
CO= 47O ILF

'#.

1k

L

I

0.1

.2
c
0

.

~

\

1

\

~

::t:

"I-"-

S
~

z

c+
j!:

10k 201

f=1 kHz

l-'

~V'

I

0.01

f=20Hz

0.005
0.001

f - Frequency - Hz

0.01
Po - Output Power - W

Figure 12

Figure 13

~TEXAS

INSTRUMENTS
2-36

10k 20k

1k

100

Po - Output Power - W

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

/

0.1

0.2

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
HEADPHONE AMPLIFIER
TOTAL HARMONIC DISTORTION + NOISE

at.

vs

vs

FREQUENCY

FREQUENCY

YO=1 Y
Ay=1
CI = 10 I1F
RI=RF=50kn
Co = 470 I1F
RL= 10kO

I

I
+

I

HEADPHONE AMPLIFIER
TOTAL HARMONIC DISTORTION + NOISE

at.

I
+

i

~..

0.1

.2

~

--

Av=1

J:

j

~

0.1

1..

I
z

YO=1 V
CI = 1Ol1F
RI= RF= 10kn
Co =470 I1F

I

I'

j

~

0.01

i!=
100

1k

10k 20k

~

..... v-+-

~

'-

0.01

Z

~

0.004
20

l~~

~y=

~

~

"]"

0.004
20

10k 20k

100

f - Frequency - Hz

f - Frequency - Hz

Figure 14

Figure 15
CLAS9-D AMPLIFIER
GAIN and PHASE

vs
FREQUENCY

,

10
9
Gain

8

~-

7

60°

l- 30°

II

6

I'

5

~I-

3
2

1,

YDD=5Y
f- PO=2W

10

RL=40

..

~

§
0'

Phase

4

o

90°

-30°

I

~

-80°

-90°

100

1k
f - Frequency - Hz

10k

30k

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-37

TPA005D14
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
HEAOPHONE AMPLIFIER

GAIN and PHASE

vs
FREQUENCY
3
2

1

...... ~

1SOO

-

II

.....

Gain

0

.........

-1

~

-2

c

-4

i-

-3

'a
I

ii
CI

IPhase

-5

l"-

I I

........ 1-

-6 f- VOO=5V
PO=40mW
-7 r-- AV=1
-6 f- CI=1OI1F
RI= RF= 10 kn
-9 r-- Co = 470 j1F
-10
100
20

-1200

-1800
1k

10k

30k

f - Frequency - Hz

Figure 17
CLAS8-0 AMPLIFIER

POWER DISSIPATION

CROSSTALK

va

vs

FREQUENCY
-36

OUTPUT POWER
3.0 , - - - - - , . - - - , - - - . . , . . - - - - , - - - ,

IV~~I~~IV

Class-O AmplHler

PO=2W
I- RL=4n

ID

'a
I

I

/

-44

~
I

2.5

~--+---I----+--F-l---I

2.0

~~-+---I--_____.ff-------1---I

j

)

-48

1.5 ~--+----f->j~'----+-------1---I

~

lii 1.0 ~--t~'f_-I----r-----j~--I

-52

-56

!.

~

-

~

/~

0.5

-60
20

100

1k

10k 20k

~~~----+---+-----+----1

O~-~--~--~-~~-~
1.0
1.5
2.0
2.5
0.5

o

f - Frequency - Hz

Po - Output Power - W

Figure 18

Figure 19

~1ExAs

2-38

INSTRUMENTS
POST OFACE 80X 855303 • DAUAS. TEXAS 75265

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
EFFICIENCY
VB

OUTPUT POWER

90
Class-D Amplifier

85

I

RL=8O

-/
75

80
'#.
I

I

70

ffi

iI'"

IV
II
/1
,/

~
c 6S

.~

60

55
50

"",.,

" . ""--;;L=40

-

4S
40

o

0.5

1.0

1.5

2.0

2.5

Po - Output Power - W

Figure 20

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-39

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, C,
In the typical application 'an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and R'N, the TPAOOS014's input resistance forms a
high-pass filter with the corner frequency determined in equation 1.

(1)

fC(highpass) = 211:i l C I

z, is nominally 10 kO
The value of C, is important to consider as it directly affects the bass (lOW frequency) performance of the circuit.
Consider the example where the specification calls for a flat bass response down to 40 Hz. Equation 1 is
reconfigured as equation 2.

CI

= _1_

(2)

211:Zlfc

In this example, C, is 0.40 I1F so one would likely choose a value in the range of 0.4711F to 1 I1F. A low-leakage
tantalum or ceramic capacitor is the best choice for the input capacitors. When polarized capacitors are used,
the positive side of the capacitor should face the amplifier input as the dc level there is held at 1.S V, which is
likely higher than the source dc level. Please note that it is important to confirm the capacitor polarity in the
application.

differential input
The TPAOOS014 has differential inputs to minimize distortion at the input to the IC. Since these inputs nominally
sit at 1.S V, dc-blocking capacitors are required on each of the four input terminals. If the signal source is
single-ended, optimal performance is achieved by treating the signal ground as a signal. In other words,
reference the signal ground at the signal source, and run a trace to the dc-blocking capacitor which should be
located physically close to the TPAOOS014. If this is not feasible, it is still necessary to locally ground the unused
input terminal through a dc-blocking capacitor.

power supply decoupling, Cs
The TPAOOS014 is a high-performance Class-O CMOS audio amplifier that requires adequate power supply
decoupling to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling
also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling
is achieved by using two capacitors of different types that target different types of noise on the power supply
leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-seriesresistance (ESR) ceramic capacitor, typically 0.1 I1F placed as close as possible to the device's various Voo
leads works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 I1F
or greater placed near the audio power amplifier is recommended.
The TPAOOS014 has several different power supply terminals. This was done to isolate the noise resulting from
high-current switching from the sensitive analog 'circuitry inside the IC.

~TEXAS

2-40

INSTRUMENTS
POST OFFICE BOX 655303 • DAu.AS, TEXAS 75265

TPA005D14
2·W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
mute and shutdown modes
The TPA005D14 employs both a mute and a shutdown mode of operation designed to reduce supply current,
100, to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN
input terminal should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN
low causes the outputs to mute and the amplifier to enter a low-current state, 100 = 0.211A. Mute mode alone
reduces 100 to 10 mAo

using low-ESR capacitors
Low-ESR capaCitors are recommended throughout this applications section. A real (as opposed to ideal)
capaCitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

output filter components
The output inductors are key elements in the performance of the class-D audio amplifier system. It is important
that these inductors have a high enough current rating and a relatively constant inductance over frequency and
temperature. The current rating should be higher than the expected maximum current to avoid magnetically
saturating the inductor. When saturation occurs, the inductor loses its functionality and looks like a short circuit
to the PWM signal, which increases the harmonic distortion considerably.
A shielded inductor may be required if the class-D amplifier is placed in an EMI sensitive system; however, the
switching frequency is low for EMI considerations and should not be an issue in most systems. The dc series
resistance of the inductor should be low to minimize losses due to power dissipation in the inductor, which
reduces the efficiency of the circuit.
Capacitors are important in attenuating the switching frequency and high frequency noise, and in supplying
some of the current to the load. It is best to use capacitors with low equivalent-series-resistance (ESR). A low
ESR means that less power is dissipated in the capaCitor as it shunts the high-frequency signals. Placing these
capaCitors in parallel also parallels their ESR, effectively reducing the overall ESR value. The voltage rating is
also important, and, as a rule of thumb, should be 2 to 3 times the maximum rms voltage expected to allow for
high peak voltages and transient spikes. These output filter capacitors should be stable over temperature since
large currents flow through them.
For a-n loads, double the inductor value and halve the common-mode capaCitors (i.e., 15 IlH to 30 IlH). For
more information, see application report SLOA023, Reducing and Eliminating the Class-D Output Filter and
application report SLOA031, Design Considerations for Class-D Audio Power Amplifiers.

~TEXAS

INSTRUMENTS
POST OFRCE BOX 655303 • DALLAS, TEXAS 75265

2-41

TPA005D14
2·W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS240A- AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
efficiency of class-D vs linear operation
Amplifier efficiency is defined as the ratio of output power delivered to the load to power drawn from the supply.
In the efficiency equation below, PL is power across the load and Psup is the supply power.

P
Efficiency = 11 = _L_
P suP

A high-efficiency amplifier has a number of advantages over one with lower efficiency. One of these advantages
is a lower power requirement for a given output, which translates into less waste heat that must be removed
from the device, smaller power supply required, and increased battery life.
Audio power amplifier systems have traditionally used linear amplifiers, which are well known for being
inefficient. Class-D amplifiers were developed as a means to increase the efficiency of audio power amplifier
systems.
A linear amplifier is deSigned to act as a variable resistor network between the power supply and the load. The
transistors operate in their linear region and voltage that is dropped across the transistors (in their role as
variable resistors) is lost as heat, particularly in the output transistors.
The output transistors of a class-D amplifier switch from full OFF to full ON (saturated) and then back again,
spending very little time in the linear region in between. As a result, very little power is lost to heat because the
transistors are not operated in their linear region. If the transistors have a low ON resistance, little voltage is
dropped across them, further reducing losses. The ideal class-D amplifier is 100% efficient, which assumes that
both the ON resistance (rOS(ON» and the switching times of the output transistors aTe zero.
the Ideal class-O amplifier

To illustrate how the output transistors of a class-D amplifier operate, a half-bridge application is examined first
(Figure 21).
.

VDD

J

~
l

J

I~
+

Rl
clI

c

vOUT

r

-::-

Figure 21. Half-Bridge Class-D Output Stage

Figures 22 and 23 show the currents and voltages of the half-bridge circuit. When transistor M1 is on and M2
is off, the inductor current is approximately equal to the supply current. When M2 switches on and M1 switches
off, the supply current drops to zero, but the inductor keeps the inductor current from dropping. The additional
inductor current is flowing through M2 from ground. This means that VA (the voltage at the drain of M2, as shown
in Figure 21) transitions between the supply voltage and slightly below ground. The inductor and capacitor form
a low-pass filter, which makes the output current equal to the average of the inductor current. The low pass fiHer
averages VA, which makes VOUT equal to the supply voltage multiplied by the duty cycle.

~TEXAS

2-42

INSTRUMENTS
POST OFFICE BOX 65S303 • DAUAS. TEXAS 75265

TPA005D14
2·W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

AppLICATiON iNfORMATiON
the Ideal class-D amplifier (continued)
Control logic is used to adjust the output power, and both transistors are never on at the same time. If the output
voltage is rising, M1 is on for a longer period of time than M2.
Inductor Current

o~---+--~--~----~--~--~--~--~----~

M1 onl M1 Off l M1 onl
M20ffl M2 on 1M2 offl •••

Time

Figure 22. Class-D Currents

~--~--~--~---'----r---'---~--~----VDD

VOUT

M10n IM1 off IM1 onl
M20ff IM20n IM20ffl···
Time

Figure 23. Class-D Voltages

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-43

TPA005D14
2·W.STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
the ideal class·O amplifier (continued)
Given these plots, the efficiency of the class-O device can be calculated and compared to an ideal linear
amplifier device. In the derivation below, a sine wave of peak voltage (Vp) is the output from an ideal class-O
and linear amplifier and the efficiency is calculated.

CLASS·O

LINEAR

Vp

V L(rms)
A

Vp

= .f2

I)
verage (00

VL(rms) =

=

IL(rms) x VL(rms)
V
00

P

.f2

_ V L(rms)2 = V p 2

L -

RL

2 RL

Average (100)
Psup = Voo x Average(l oo)

P

Voox IL(rms) x VL(rms)
Voo

- ------';,..--'-------'-----'-

SUP -

Efficiency

Efficiency

=

=

YJ

YJ

PL

=P
sup

=1

Psup

= Voo

=~ x

V

RP
L

x Average ( 100 )

=

Voo Vp 2
R
x 3t
L

PL
Efficiency = YJ = - Psup

Efficiency

=

YJ

Efficiency

=

YJ

V

= ~ x .--.E..
4

VOO

In the ideal efficiency equations, assume that Vp =Voo, which is the maximum sine wave magnitude without
clipping. Then, the highest efficiency that a linear amplifier can have without clipping is 78.5%. A class-O
amplifier, however, can ideally have an efficiency of 100% at all power levels.
The derivation above applies to an H-bridge as well as a half-bridge. An H-bridge requires approximately twice
the supply current but only requires half the supply voltage to achieve the same output power-factors that
cancel in the efficiency calculation. The H-bridge circuit is shown in Figure 24.

voo

voo
+ vOUT-

L
L

Figure 24. H·Bridge Class·O Output Stage

~1ExAS

2-44

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
losses in a real-world class-D amplifier
Losses make class-O amplifiers non ideal , and reduce the efficiency below 100%. These losses are due to the
output transistors having a nonzero r08(on), and rise and fall times that are greater than zero.
The loss due to a nonzero r08(on) is called conduction loss, and is the power lost in the output transistors at
nonswitching times, when the transistor is ON (saturated). Any R08(on) above 0 n causes conduction loss.
Figure 25 shows an H-bridge output circuit simplified for conduction loss analysis and can be used to determine
new efficiencies with conduction losses included.
VOO=5V

ROS(on)

0.35 0

5 MO

RDS(off)

0.35 0

RDS(on)

40
ROS(Off)

5 Mel

Figure 25. Output Transistor Simplification for Conduction Loss Calculation
The power supplied, P8UPo is determined to be the power outputto the load plus the power lost in the transistors,
assuming that there are always two transistors on.
PL

Efficiency = '11 -- P
8UP
Efficiency = '11

Efficiency = '11

12 2r08(on) + 12RL
RL
2r08(on)

+ RL

Efficiency = '11

= 95%

(at all output levels r08(on)

= 0.1,

Efficiency = '11

= 85%

(at all output levels r 08(on)

= 0.35,

-!I
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75285

RL

= 4)

RL

= 4)

2-45

TPAOO5D14
2-W STEREO CLAS8-D AUDIO POWER AMPLIFIER
SL0S240A- AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
losses In a real-world class-D amplifier (continued)

Losses due to rise and fall times are called switching losses. A plot of the output, showing switching losses, is
shown in Figure 26.

HtsWon

+

H
tswoff

= tsw

Figure 26. Output SWitching Losses

Rise and fall times are greater than zero for several reasons. One is that the output transistors cannot switch
instantaneously because (assuming a MOSFET) the channel from drain to source requires a specific period
of time to form. Another is that transistor gate-source capacitance and parasitic resistance in traces form RC
time constants that also increase rise and fall times.
Switching losses are constant at all output power levels, which means that switching losses can be ignored at
high power levels in most cases. At low power levels, however, switching losses must be taken into account
when calculating efficiency. Switohing losses are dominated by conduction losses at the high output powers,
but should be considered at low powers. The switching losses are automatically taken into account if you
consider the quiescent current with the output filter and load.
class-D effect on power supply

Efficiency calculations are an important factor for proper power supply design in amplifier systems. Table 2
shows class-D efficiency at a range of output power levels (per channel) with a 1-kHz sine wave input. The
maximum power supply draw from a stereo 1-W per channel audio system with s-n loads and a 5-V supply is
almost 2.7 W. A similar linear amplifier such as the TPAOO5D14 has a maximum draw of 3.25 W under the same
circumstances.
Table 2. Efficiency vs Output Power in 5-V &-n H-Bridge Systems
output Power (W)

EffIciency (%)

Peak Voltage (V)

Internal DIssIpation (W)

0.25
0.5

63.4
73
77.1

2
2.83
3.46

0.145
0.183
0.222

4
4.47t

0.314
0.3

0.75
1
79.3
1.25
80.6
t High peak voltages cause the THO to increase

~TEXAS

2-46

INSTRUMENTS
POST OFFICE BOX II55S03 • DALLAS, l£XAS 75265

TPAOOSD14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
class-D effect on power supply (continued)
There is a minor power supply savings with a class-O amplifier versus a linear amplifier when amplifying sine
waves. The difference is much larger when the amplifier is used strictly for music. This is because music has
much lower RMS output power levels, given the same peak output power (Figure 27); and although linear
devices are relatively efficient at high RMS output levels, they are very ineffiCient at mid-to-Iow RMS power
levels. The standard method of comparing the peak power to RMS powerfor a given signal is crestfactor, whose
equation is shown below. The lower RMS power for a set peak power results in a higher crest factor
Crest Factor = 10 log

PPK

P nns

Time

Figure 27. Audio Signal Showing Peak and RMS Power
Figure 28 is a comparison of a 5-V class-O amplifier to a similar linear amplifier playing music that has a 13.76-dB
crest factor. From the plot, the power supply draw from a stereo amplifier that is playing music with a 13.76 dB·
crest factor is 1.02 W, while a class-O amplifier draws 420 mW under the same conditions. This means that just
under 2.5 times the power supply is required for a linear amplifier over a class-O amplifier.
POWER SUPPLIED

vs
PEAK OUTPUT VOLTAGE AND PEAK OUTPUT POWER

600

500

I

400

".!!

I

~

III

J

TPA0202
300

200

~

~

......

.".......

~

~

TPA005D14

-~

100

o
1
0.25

/

1.5
0.56

2

2.5
1.56

3
2.25

V

--

-

3.5

4

3.06

4

4.5
5.06

Peak Output Voltage (V)
Peak Output Power (W)

Figure 28. Audio Signal Showing Peak and RMS Power (With Music Applied)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 656303 • DALlAS, TEXAS 75265

2-47

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
class-D effect on battery life
Battery operations for class-D amplifiers versus linear amplifiers have similar power supply savings results. The
essential contributing factor to longer battery life is lower RMS supply current. Figure 29 compares the
TPA005D14 supply current to the supply current of the TPA0202, a 2-W linear device, while playing music at
different peak voltage levels.
SUPPLY CURRENTS

vs
PEAK OUTPUT VOLTAGE AND PEAK OUTPUT POWER
400
350
'il'

E

300

V

c(

.s

250

~

200

c
::s

TPA0202/

0

~

a.
a.
::s
U)

./

150
100
50

~

~

V"
TPAO~5D.:!!- ~

~

o
1

0.25

1.5
0.56

2
1

2.5
1.56

3
2.25

3.5
3.06

4
4

Peak Output Voltage (V)
Peak Output Power (W)

Figure 29. Supply Current vs Peak Output Voltage of TPA005D14 vs TPA0202 With Music Input
This plot shows that a linear amplifier has approximately three times more current draw at normal listening levels
than a class-D amplifier. Thus, a class-D amplifier has approximately three times longer battery life at normal
listening levels. If there is other circuitry in the system drawing supply current, that must also be taken into
account when estimating battery life savings.

~TEXAS

INSTRUMENTS
2--48

POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

TPAOOSD14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA005D14 data sheet, one can see that when the
TPA005D14 is operating from a 5-V supply into a 4-Q speaker that 4 W peaks are available. Converting Watts
to dB:

(1)

= 6 dB
P dB = 10Log (P w) = 10Log
P ref
1

(3)

Subtracting the crest factor restriction to obtain the average listening level without distortion yields:
6.0 dB - 18 dB

- 12 dB (15 dB crest factor)

6.0 dB - 15 dB = - 9 dB (15 dB crest factor)
6.0 dB - 12 dB = - 6 dB (12 dB crest factor)
6.0 dB - 9 dB = - 3 dB (9 dB crest factor)
6.0 dB - 6 dB = - 0 dB (6 dB crest factor)
6.0 dB - 3 dB = 3 dB (3 dB crest factor)
Converting dB back into watts:

Pw

=

10PdB/10 x P ref

(4)

= 63 mW (18 dB crest factor)

=

125 mW (15 dB crest factor)

= 250 mW (12 dB crest factor)

= 500 mW (9 dB crest factor)

=

1000 mW (6 dB crest factor)

= 2000 mW (3 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 4-Q system, the internal dissipation in the TPAOO5D14
and maximum ambient temperatures is shown in Table 3.

~1ExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

2-49

TPAOO5D14
2·W STEREO CLAS8-D AUDIO POWER AMPLIFIER
SL0S240A - AUGUST 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations (continued)
Table 3. TPAOO5D14 Power Rating, S-V, 4-0, Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT
POWER

POWER DISSlPAnoN
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

4

2W(3dB)

0.56

4

1000 mW (6 dB)

0.30

4

500mW(9dB)

0.23

139"Ct

4

250 mW (12 dB)

0.20

141°Ct

4

120 mW (15 dB)

0.14

143°Ct

0.09

14soCt

63 mW (18 dB)
4
t Case temperature (TC) IS rated to 125°C maximum.

' 125°C
136°Ct

DlSSIPAnON RAnNG TABLE
PACKAGE

TA s 25"C

DERAnNG FACTOR

TA =70°C

DCA

5.6 W

44.8 mwrc

3.5 W

2.9W

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
data from the dissipation rating table, the derating factor for the DCA package with 6.9 in2 of copper area on
a multilayer PCB is 44.8 mWfOC. Converting this to 9JA:

~

1

~-

~

=_1_
0.0448
= 22.3°CfW

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given 9JA, the maximum
allowable junction temperature, and the total intemal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA005D14
is 150°C. The intemal dissipation figures are taken from the Efficiency vs Output Power graphs.
TA Max = TJ Max - 9 JA Po

(6)

150 - 22.3(0.14 x 2)

143°C (15 dB crest factor)

150 - 22.3(0.56 x 2)

125°C (3dB crest factor)
NOTE:

Internal dissipation of 0.6 W is estimated for a 2-W system with a 15 dB crest factor per channel.

Table 3 shows that for some applications no airflow is required to keep junction temperatures in the specified
range. The TPA005D14 is designed with thermal protection that tums the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Table 3 was calculated for maximum listening
volume without distortion. When the output level is reduced the numbers in the table change Significantly. Also,
using 8-0 speakers dramatically increases the thermal performance by increasing amplifier efficiency.

~TEXAS

2-50

INSTRUMENTS
POST OFFICE BOX 65/i303 • DAllAS. TEXAS 75265

TPA005D14
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS240A - AUGUST 1999 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced DCA package is based on the 56-pin TSSOP, but includes a thermal pad (see Figure 30)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an extemal heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Thermal
Pad

(~

I

DIE

~ EHj E1 E1 rl

Side VIew <~~

,~~
~ III/);

i~
..

I

-=-

lCOMP

I

PVDD

vDDIL- VDD

BIAS
GENERATOR

I
I

RCOMP

I

• 1:/
•

TRIPlER
CHARGE PUMP

1.5V

1- ......~

..... Ih
~
L---

RPVDD

______ _____

PGND

__________

.

_i!I
~•

________ J

i!I

~

!:jO

~

C

~

~
~

."

t:
::!!

!II

VCP-UVlO
DETECT
PVDD

I
I
I
I
RPVDD

_DO

lI

RlNP
RINNI

-=-

-=-

RAMP

10

:D
V2P5

coscH
I""""'" I

I

~

lPVDD

~

Ii o
i »~

m~

5

GATE
DRIVE

~&:

--"<:-() () ()
s;) J ;S ~

ZO

NOTE B. lPVDD. RPVDD. VDD. and PVDD are externally connected. AGND and PGND are externally connected.

~

TPAOOSD02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S227C - AUGUST 1998 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME

AGND

DESCRIPTION

NO.
3.7,20,
46,47

Analog ground for analog sections

COSC

46

Capacitor 1/0 for ramp generator. Adjust the capacitor size to change the switching frequency.

CP1

25

First diode node for charge pump

CP2

24

First inverter switching node for charge pump

CP3

23

Second diode node for charge pump

CP4

26

Second inverter switching node for charge pump

FAULTO

42

Logic level faullO output signal. Lower order bit of the two fault signals with open drain output.

FAULT1

41

Logic level fault1 output signal. Higher order bit of the' two' fault signals with open drain output.

LCOMP

6

Compensation capacitor terminal for left-channel Class-D amplifier

LINN

4

Class·D left·channel negative input

L1NP

'5

Class-D left-channel positive input

LOUTN

14,15

LOUTP

10,11

Class-D amplifier left-channel positive output of H-bridge

LPVDD
LSBIAS

9,16

Class-D amplifier left-channel power supply

MUTE

NC

Class-D amplifier left-channel negative output of H-bridge

28

Level-shifter power supply, to be tied to VCP

2

Active-low logic-level mute input signal. When MUTE is held low, the selected amplifier is muted. When MUTE
is held high, the device operates normally. When the Class-Damplifier is muted, the low-side output transistors
are tumed on, shorting the load to ground.

17,18,19,
30,31

PGND

12,13

PGND

27

PGND

36,37

PVDD
RCOMP

21,32

No intemal connection
Power ground for left-channel H-bridge only
Power ground for charge pump only
Power ground for right-channel H-bridge only
VDD supply for charge-pump and intemallogic circuitry

43

Compensation capacitor terminal for right-channel Class-D amplifier

RINN

45

Class-D right-channel negative input

RINP

Class-D right-channel positive input

RPVDD

44
33,40

ROUTN

34,35

Class-D amplifier right-channel negative output of H-bridge

ROUTP

38,39

SHUTDOWN

1

Class-D amplifier right-channel power supply
Class-D amplifier right-channel positive output of H-bridge
Active-low logic-level shutdown input signal. When SHUTDOWN is held low, the device goes into shutdown
mode. When SHUTDOWN is held at logic high, the device operates normally.

V2P5

29

2.5-V intemal reference bypass

VCP

22

Storage capacitor terminal for charge pump

VDD

8

VDD bias supply for analog circuitry. This terminal needs to be well filtered to prevent degrading the device
performance.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655300 • DALlAS, TEXAS 75265

2-55

TPA005D02
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

Class-D amplifier faults
Table 1. Amplifier Fault Table

t

FAULTot

FAULT1t

1

1

No fault-The device is operating normally.

DESCRIPTION

1

0

Charge pump under-voltage lock-out (VCP-UV) fault-All low-side transistors are turned on, shorting the load to
ground. Once the charge pump voltage is rastorad, normal operation resumes, but FAULTl is still active. FAULT1 is
cleared by cycling MUTE, SHUTDOWN, or the power supply.

0

0

Thermal fault-All the low-side transistors are turned on, shorting the load to ground. Once the junction temperature
drops 20°C, normal operation resumes. But the FAULTx terminals are still set and are cleared by cycling MUTE,
SHUTDOWN, or the power supply.

These logic levels assume a pull up to PVDD from the open-drain outputs.

AVAILABLE OPTIONS
PACKAGED DEVICES

t

TA

TSSOpt
(DCA)

-40°C to 125°C

TPAOO5D02DCA

The DCA package IS available In left-ended tape and reel. To order
a taped and reeled part, add the suffix R to the part number (e.g.,
TPA005D02DCAR).

=

absolute maximum ratings over operating free-air temperature range, TC 25°C (unless otherwise
noted)*
Supply voltage, VDO (PVoo, LPVoo, RPVoo, Voo) ........................................... 5.5 V
Bias voltage (LSBIAS) .............................................................. 12 V to 20 V
Input voltage, VI (SHUTDOWN, MUTE, MODE) ...................................... -0.3 V to 5.8 V
Output current, 10 (FAULTO, FAULT1), open drain terminated ................................... 1 mA
Charge pump voltage, VCP .......................................................... PVoo + 20 V
Continuous H-bridge output current .......................................................... 2 A
Pulsed H-Bridge output current, each output, Imax (see Note 1) .................................. 5 A
Continuous total power dissipation, TC = 25°C .............................................. 4.5 W§
Operating virtual junction temperature range, TJ .................................... -40°C to 150°C
Operating case temperature range, TC ............................................ -40°C to 125°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
t Stresses beyond those listed under "absolute maximum ratings· may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicaied under "recommended operating conditions· is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
§ Thermal shutdown activates when TJ = 125°C.
NOTE 1: Pulse duration = 10 ms, duty cycle s 2%
DISSIPATION RATING TABLE
PACKAGE

TAS25°C~
POWER RATING

DCA

5.6W

DERATING FACTOR
ABOVE TA = 25°C

=

=

=

TA 70°C
POWER RATING

TA 85°C
POWER RATING

TA 125°C
POWER RATING

3.6W

2.9W

1.lW

11 Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Repott(literature number
SLMAOO2), for more information on the PowerPAD package. The thermal data was measured on a PCB layout based on the
information in the section entitled Texas Instruments Recommended Board for PowerPADon page 33 of the before mentioned
document.

~TEXAS

INSTRUMENTS
2-56

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA005D02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 199B - REVISED MARCH 2000

recommended operating conditions
MIN

NOM

4.5

Supply voltage, PVOD, LPVOO, RPVOO, VOO
High-level input voltage, VIH

V

0.75

V

V

Audio inputs, LINN, L1NP, RINN, RINP, HPLlN, HPRIN, differential input voltage

1

PWM frequency

100

electrical characteristics, Voo = PVoo = LPVoo = RPVoo = 5 V, RL = 4 n, Tc
(resistive load) (unless otherwise noted)
PARAMETER

TEST CONDITIONS

500

MIN

TYP

Power supply rejection ratio

VOO = PVOO = LPVOO = RPVOO = 4.9 V to 5.1 V

40

100

Supply current

No load or output filter

25

IOO(MUTE)

Supply current, mute mode

MUTE=OV

IOO(SO)

Supply current, shutdown mode

SHUTOOWN = 0 V

IIH

High-level input current

VIH=5.3V

IlL

Low-level input current

VIL=-0.3V

rOS(on)

Total static drain-to-source
on-state resistance
(low-side plus high-side FETs)

IO=0.5A

rOS(on)

Matching

MAX

UNIT
dB
mA

10

15

mA

400

2000

-10

ItA
ItA
ItA

750

mO

620

TEST CONDITIONS

kHZ

40

10

95%

PARAMETER

VRMS

=25°C, See Figure 1

PSRR

operating characteristics, Voo = PVoo = LPVoo = RPVoo = 5 V, RL = 4 n, Tc
(unless otherwise noted)

UNIT

5.5

4.25

Low-level input voltage, VIL

99.5%

= 25°C, See Figure 1

MIN

TYP

Po

RMS output power, THO = 0.5%, per channel

THO+N

Total harmonic distortion plus noise

PO=IW, f= 1 kHz

0.2%

Efficiency

RL=80

80%

AV

Gain

MAX

2

UNIT
W

24

Left/right channel gain matching

dB

95%

Noise floor

60

dB

Dynamic range

70

dB

Crosstalk

55

f = 1 kHz

Frequency response bandwidth, post output filter, -3 dB
BOM

MAX

20

Maximum output power bandwidth

dB
20,000

Hz

20

kHz

thermal resistance

I TEST CONDITIONS

PARAMETER
RaJp

Thermal resistance, junction-to-pad

RaJA

Thermal resistance, junction-to-padt

MIN

TYP

I
22.3

MAX

UNIT

10

°C/W
°C/W

t Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report (literature number SLMA002), for
more information on the PowerPAO package. The thermal data was measured on a PCB layout based on the information in the section entitled
Texas Instruments Recommended Board for PowerPAD on page 33 of the before mentioned document.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

2-57

TPAOOSD02
2·W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S227C - AUGUST 1998 - REViseD MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r-------------------,

1

FAULTOU!141
FAULT1 t-"-'-

1

VCP ~ LSBIAS
PVDD
PVDD

~

1
1 14,15

2 1 SHUTDOWN
MUTE

--e___-__,

LOUTN~:<.:..::...JYYY\___

~

1
1

5V 9•16

LPVDD

1

111F

Balanced
Differential
Input Signal

I
1

{---1f-ii
~~

UNP

1'1 UNN
111F

-.Cl

61 LCOMP

470PF~

J

470 PFT,

-=-

I

RCOMP

1

r l!

cosc

470 PFT

1

-=111F
Balanced
Differential
Input Signal

I

{---1~

~~

l
r-=_T-I

;-=_--,t

1 \ 1 RINN
111F

5V
33,40
2372046 7
1213273637

5V

1

I

RPVDD
AGND <_Note A)
PGND <_ Note A)

l

T

47 nF

2.211F

1
1
21,32 PVDD
1

1
1
1
1
1
1

ROUTP~~~~~~~~--*-~

1

L ___________________~
1
1

Figure 1. 5-V, 4-0 Test Circuit

2-58

47 nF

RINP

-!i11ExAs
INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

TPA005D02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Switching frequency

Supply current

100

Total harmonic distortion plus noise

THO+N

2

vs Free-air temperature

3

vs Frequency

4,5

vs Output power

6,7

Voltage amplification and phase shift

vs Frequency

Crosstalk

vs Frequency

9

Efficiency

vs Output power

10

8

SUPPLY CURRENT -

SUPPLY CURRENT

vs

vs

SWITCHING FREQUENCY

FREE-AIR TEMPERATURE

50

100

Open Load

Open Load

C
E

\

80

60

(J

Do
:;,
Ul
I

40

I

Y
- -----

C

i
a

C

E

I

C
~
:;,
(J

With Output Filter

40

~

~

.......

0
_0

L..---" ~

20

With Output Riter

30

Do
Do
:;,
Ul
I

20

'\

0
_0

Without Output Filter

10

\ i t h o u t Output Filter

o

o

100

200

300

400

o
500

-40

o

40

80

120

125

TA - Free-Air Temperature - 'C

f - Switching Frequency - Hz

Figure 2

Figure 3

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2-59

TPA005D02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY
~
I

RL=40
Wll

VV
1/1/

11111

-1I~~=2W ~

0.2
0.1

\1"'1
1\
f::
r-

~
L

~

"

Po=100 mW

RL=80
0.5

,

o

0.2

i

0.1

i!
.~

Is

=t=I= PO=500mW

.A

lc

~

/

1".1

PO=1W

0.05

I

./

I

Z

0.01
20

,~

i\

~
.i

-

0.05

so 100 200

500 1k

2k

5k 10k 20k

~

0.02
0.01
20

so 100 200

SOO 1k

TOTAL HARMONIC DISTORTION PLUS NOISE.

vs

OUTPUT POWER

OUTPUT POWER

10

5 ~ RL=40

5

2

2

-

t-tlllill

r......

I

1= 1 kHz
~

I,

0.2

...r'

"t-

f=2OkHz .

f=20Hz

0.1

~f=20kHz

0.05 t----+--

0.05 r-- r-

r-- r-

1= 20 Hz

·1

0.02
0.01
0.01 0.02

RL=80

0.5
f=1kHz

..... Iiiii;;

I 1111

I III I
0.05 0.1 0.2

5k 10k 20k

Figure 5

vs
10

2k

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

0.5

1

2
Po - output Power - W

5

10

.1

r~2O~Z
f=20kHz

0.02
0.01
10m 20m 50m 100m 200m SOOm 1

2

Po - Output Power - W

Figure 6

Figure 7

~TEXAS

INSTRUMENTS
2-60

/

V"

PO=1W

Figure 4

0.1

/ /:::=
i"""

::::.....

f - Frequency - Hz

0.2

V

PO=100mW

~

0.02

0.5

!

POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

5

10

TPA005D02
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S227C - AUGUST 1998 - REVISED MARCH 2000

TYpiCAL CHARACTERISTiCS
GAIN AND PHASE

vs
FREQUENCY

30
28
26 24
22
20
18
16
14
12
10
8
6
4

III
'1:1

I

c
iii

CJ

11111

I"""I

Po=2W
RL=4ll

~~Wag~ A~~IINiiatl~n

"
Phase Shift

2

45°
40°
35°
30°
25°
20°
15°
10°
5°
0°
-5°
-10°
-15°
-20°
-25°
-30°
-35°

J
Q.

_~

o

10 20

50 100200 500 lk 2k

-45°
5k10k2Ok50kl00k

f - Frequency - Hz

FigureS
CROSSTALK

EFFICIENCY

vs

vs

FREQUENCY

OUTPUT POWER

0
-10

90

-20
-30
III
'1:1

I

1e
(J

-80

-90
-100
-110
-120
-130
-140
-150
20

RL=8n~

"",

80

-40
-60

-60
-70

I

PO=2W
RL=4ll

..,.
~

Left-to-Rlght

\ \ jill

I

ffi

I IIII

Right-to-Left

60

50

50 100 200

500 lk

2k

5k 10k 20k

V/ \

(JV

~
c
.!!!

.!!

_\J

/

70

RL=4ll

~

I

40 0

0.4

0.8

1.2

1.6

2.0

Po - Output Power - W

f - Frequency - Hz

Figure 9

Figure 10

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TPA005D02
.
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced DCA package is based on the 56-pin TSSOP, but includes a thermal pad (see Figure 11)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today'sadvanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Side View (a)

Thermal
Pad

End View (b)

BoHom View (c)

Figure 11. Views of Thermally Enhanced DCA. Package

selection of components
Figure 12 is a schematic diagram of a typical notebook computer application circuit.

~lExAs

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TPA005D02
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

r-------------------,

FAULTO~

I

I

I 41

YCP ~ LSBIAS

FAULTl

--.!JI
PYDD

SHUTDOWN
PVDD ~ MUTE

1511H
LOUTN 1:--'-'''-''"'-fY'''"''"'--.---4It-----,

I
I

Balanced
Differential
Input Signal

5 y 9•16

I

ll1F

I
I

--1~1
ll1F

tl

470 pF-.L
T

-=-

I
I
LPYDD

470 PFT

1

LINN

V2P5

I

RCOMP

1

I
I

I

I

I

CPl 11-'2=5_--,

1

47nF
124
CP21
CP3 11-'2=3_--"
I
T"-'--'47nF
CP41;-2",6'-----'-

f

{ ~~

J~ RINN
1 I1F I

.~I

I

5V
I RPYDD
2.3.7,20,46,47 I AGND (see Note A)
112,13,27,36,37 PGND (see Note A)

l

YCP 11-'2",2,-----,
I
I
T

i
I
I

5Y

ll1F

I
I
I

--1~ RINP

J-

I

Is
VDDr-

t

ll1F

Balanced
Differential
Input Signal

In9

LCOMP

I

-=-

40

LOUTP 1011
I

-=- I
~ cosc
~
!

470PFT

ll1F

1

{ --1~ LlNP

r -_ _ _....::6'-11

r-=-

I11415

2.211F

I
I

2~ PVDD

I

ROUTN~M~3~5~yy~.---~-___,

I

I

I

I

I

I
I

I
I

0.2211Fh

I
0.2211F
ROUTPI-'~~3~9-Fnn~~-~~-~

I
I

IL _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ I
~

NOTE A. A O.1I1F ceramic capacitor should be placed as close as possible to the Ie. For filtering lower-frequency noise signals, a larger aluminum
electrolytic capacitor of 10 I1F or greater should be placed near the audio power amplifier.

Figure 12. TPA005D02 Typical Configuration Application Circuit

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TPAOO5D02
2·W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RIN, the TPAOOSD002's input resistance forms a
high-pass filter with the comer frequency determined in equation 8.

(8)

fC(highpass)

RIN is nominally 10 kn
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where the specification calls for a flat bass response down to 40 Hz. Equation 8 is
reconfigured as equation 9.

C,

=

1
2ltRINf C

(9)

In this example, CI is 0.40 JlF so one would likely choose a value in the range of 0.47 JlF to 1 JlF. A low-leakage
tantalum or ceramic capacitor is the best choice for the input capacitors. When polarized capacitors are used,
the positive side of the capacitor should face the amplifier input as the dc level there is held at 1.S V, which is
likely higher than the source dc level. Please note that it is important to confirm the capacitor polarity in the
application.

differential input
The TPAOOSD02 has differential inputs to minimize distortion at the input to the IC. Since these inputs nominally
sit at 1.S V, dc-blocking capacitors are required on each of the four input terminals. If the signal source is
single-ended, optimal performance is achieved by treating the signal ground as a signal. In other words,
reference the signal ground at the signal source, and run a trace to the dc-blocking capacitor which should be
located physically close to the TPA005D02. If this is not feasible, it is still necessary to locally ground the unused
input terminal through a dc-blocking capacitor.

power supply decoupling, Cs
The TPAOOSD02 is a high-performance Class-D CMOS audio amplifier that requires adequate power supply
decoupling to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling
also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling
is achieved by using two capacitors of different types that target different types of noise on the power supply
leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-seriesresistance (ESR) ceramic capacitor, typically 0.1 JlF placed as close as possible to the device's various Voo
leads works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capaCitor of 10 JlF
or greater placed near the audio power amplifier is recommended.
The TPAOOSD02 has several different power supply terminals. This was done to isolate the noise resulting from
high-current switching from the sensitive analog circuitry inside the IC.

~TEXAS

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INSTRUMENTS
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TPA005D02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

mute and shutdown modes
The TPA005D02 employs both a mute and a shutdown mode of operation designed to reduce supply current,
100, to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN
input terminal should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN
low causes the outputs to mute and the amplifier to enter a low-current state, IDO = 400!lA. Mute mode alone
reduces 100 to 10 rnA.

Table 2. Shutdown and Mute Mode Functions
OUTPUT

INPUTSt

AMPUFIER STATE

SE/BTL

HP/UNE

MUTE IN

SHUTDOWN

MUTE OUT

INPUT

Low

Low

Low

Low

Low

UR Line

BTL

X

X

-

High

-

X

Mute

OUTPUT

X

X

High

-

High

X

Mute

Low

High

Low

Low

Low

URHP

BTL

High

Low

Low

Low

Low

UR Line

SE

High

High

Low

Low

Low

URHP

SE

t Inputs should never be left unconnected.
X

=do not care

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capaCitor in the Circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

output filter components
The output inductors are key elements in the performance of the class D audio amplifier system. It is important
that these inductors have a high enough current rating and a relatively constant inductance over frequency and
temperature. The current rating should be higher than the expected maximum current to avoid magnetically
saturating the inductor. When saturation occurs, the inductor loses its functionality and looks like a short circuit
to the PWM Signal, which increases the harmonic distortion considerably.
A shielded inductor may be required if the class D amplifier is placed in an EMI sensitive system; however, the
switching frequency is low for EMI considerations and should not be an issue in most systems. The DC series
resistance of the inductor should be low to minimize losses due to power dissipation in the inductor, which
reduces the efficiency of the circuit.
Capacitors are important in attenuating the switching frequency and high frequency noise, and in supplying
some of the current to the load. It is best to use capacitors with low equivalent-series-resistance (ESR). A low
ESR means that less power is dissipated in the capacitor as it shunts the high-frequency signals. Placing these
capacitors in parallel also parallels their ESR, effectively reducing the overall ESR value. The voltage rating is
also important, and, as a rule of thumb, should be 2 to 3 times the maximum rms voltage expected to allow for
high peak voltages and transient spikes. These output filter capaCitors should be stable over temperature since
large currents flow through them.

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TPA005D02
2·W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

efficiency of class D vs linear operation
Amplifier efficiency is defined as the ratio of output power delivered to the load to power drawn from the supply.
In the efficiency equation below, PL is power across the load and Psup is the supply power.
Efficiency

P
Psup

= 11 = __L_

A high-efficiency amplifier has a number of advantages over one with lower efficiency. One of these advantages
is a lower power requirement for a given output, which translates into less waste heat that must be removed
from the device, smaller power supply required, and increased battery life.
Audio power amplifier systems have traditionally used linear amplifiers, which are well known for being
inefficient. Class D amplifiers were developed as a means to increase the efficiency of audio power amplifier
systems.
A linear amplifier is designed to act as a variable resistor network between the power supply and the load. The
transistors operate in their linear region and voltage that is dropped across the transistors (in their role as
variable resistors) is lost as heat, particularly in the output transistors.
The output transistors of a class D amplifier switch from full OFF to full ON (saturated) and then back again,
spending very little time in the linear region in between. As a result, very little power is lost to heat because the
transistors are not operated in their linear region. If the transistors have a low ON resistance, little voltage is
dropped across them, further reducing losses. The ideal class D amplifier is 100% efficient, which assumes that
both the ON resistance (RDS(ON» and the switching times of the output transistors are zero.
the ideal class D amplifier

To illustrate how the output transistors of a class D amplifier operate, a half-bridge application is examined first
(Figure 13).
voo

I~
L

+

Figure 13. Half-Bridge Class D Output Stage

Figures 14 and 15 show the currents and voltages of the half-bridge circuit. When transistor M1 is on and M2
is off, the inductor current is approximately equal to the supply current. When M2 switches on and M1 switches
off, the supply current drops to zero, but the inductor keeps the inductor current from dropping. The additional
inductor current is flowing through M2 from ground. This means that VA (the voltage at the drain of M2, as shown
in Figure 13) transitions between the supply voltage and slightly below ground. The inductor and capacitor form
a low-pass filter, which makes the output current equal to the average of the inductor current. The low pass filter
averages VA, which makes VOUT equal to the supply voltage multiplied by the duty cycle.

~TEXAS

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TPA005D02
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S227C- AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATiON
the ideal class D amplifier (continued)

Control logic is used to adjuslthe output power, and both transistors are never on atthe same time. Ifthe output
voltage is rising, M1 is on for a longer period of time than M2.
Inductor Current
1-:::a~+-"'-=---:::.~-+-'~,--,J~..:::o.",;::----:::;oI~r::"-"'--::;;~-

Output Current

Supply Current

O-r---+--~--~~--~--~--~--------------'

M1 on, M1 off 1 M1 on,

M2 off, M2 on 1M2 off, • • •

nme
Figure 14. Class D Currents

~--~--~---r--~~--r---~---r---,-----VDD

VOllT
O~---+----~--+---~--~--~~--~--~----'

M1 on IM1 off IM1 on,
M20ff ,M20n lM2off,···

nme
Figure 15. Class D Voltages

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TPA005D02
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
the ideal class D amplifier (continued)
Given these plots, the efficiency of the class D device can be calculated and compared to an ideal linear amplifier
device. In the derivation below, a sine wave of peak voltage (Vp) is the output from an ideal class D and linear
amplifier and the efficiency is calculated.

CLASSD

V

-

L(rms) -

A

LINEAR

vP

Vp

,f2

VL(rms) =

I ) = IL(rms)Vx VL(rms)
verage (00
00

P _
L-

V

,f2

L(rms)
RL

2

V 2
= _p_
2 RL

Average (100) =

RP
L
Voo Vp 2
R
x n
L

PL
Efficiency = tJ = - Psup

Voox IL(rms) x VL(rms)
Voo

-----'-;-:---'------'-----'-

SUP -

V

P sup = Voo x Average ( 100) =

P sup = Voo x Average(loo)

P

~x

V p2
PL
Efficiency = tJ = - Psup

2RL
Efficiency = tJ =Voox--2 Vp
n x RL
11:
Vp
Efficiency = tJ = - x - 4 V DD

Efficiency = tJ = 1

In the ideal efficiency equations, assume that Vp = Voo, which is the maximum sine wave magnitude without
clipping. Then, the highest efficiency that a linear amplifier can have without clipping is 78.5%. A class 0
amplifier, however, can ideally have an efficiency of 100% at all power levels.
The derivation above applies to an H-bridge as well as a half-bridge. An H-bridge requires approximately twice .
the supply current but only requires half the supply voltage to achieve the same output power-factors that
cancel in the efficiency calculation. The H-bridge circuit is shown in Figure 16.

voo

J

voo

-4

I~

l

J

L

+ VOUT-

l

Rl
clI
I

.,,-

Figure 16. H-Brldge Class D Output Stage

~TEXAS

2-68

L

Cl

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

.,,-

TPA005D02
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
losses in a real-world class D amplifier
Losses make class D amplifiers nonideal, and reduce the efficiency below 100%. These losses are due to the
output transistors having a nonzero ROS(on), and rise and fall times that are greater than zero.
The loss due to a nonzero ROS(on) is called conduction loss, and is the power lost in the output transistors at
nonswitching times, when the transistor is ON (saturated). Any ROS(on) above 0 Q causes conduction loss.
Figure 17 shows an H-bridge output circuit simplified for conduction loss analysis and can be used to determine
new efficiencies with conduction losses included.
VOO=5V

ROS(on)

0.31 0

5 MO

ROS{off)

0.31 0

ROS{on)

Rl
40
ROS{off)

5 MO

Figure 17. Output Transistor Simplification for Conduction Loss Calculation
The power supplied, PsuP, is determined to be the power outputto the load plus the power lost in the transistors,
assuming that there are always two transistors on.

PL

Efficiency = I'J = - PsuP
Efficiency
12 2R OS (on)
Efficiency

= I'J

+

12RL

RL
2R OS (on)

+

RL

Efficiency = I'J

= 95%

(at all output levels ROS(on)

= 0.1,

Efficiency = I'J

= 87%

(at all output levels Ros(on)

= 0.31,

RL

= 4)

RL

= 4)

~TEXAS

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TPA005D02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
losses in a real-world class D amplifier (continued)

Losses due to rise and fall times are called switching losses. A plot of the output, showing switching losses, is
shown in Figure 18.

H

tswon

H

+

tswoff

=

tsw

Figure 18. Output Switching Losses

Rise and fall times are greater than zero for several reasons. One is that the output transistors cannot switch
instantaneously because (assuming a MOSFET) the channel from drain to source requires a specific period
of time to form. Another is that transistor gate-source capacitance and parasitic resistance in traces form RC
time constants that also increase rise and fall times.
Switching losses are constant at all output power levels, which means that switching losses can be ignored at
high power levels in most cases. At low power levels, however, switching losses must be taken into account
when calculating efficiency. Switching losses are dominated by conduction losses at the high output powers,
but should be considered at low powers. The switching losses are automatically taken into account if you
consider the quiescent current with the output filter and load.
class D effect on power supply

Efficiency calculations are an important factor for proper power supply design in amplifier systems. Table 2
shows Class 0 efficiency at a range of output power levels (per channel) with a 1-kHz sine wave input. The
maximum power supply draw from a stereo 1-W per channel audio system with 8-0 loads and a 5-V supply is
almost 2.7 W. A similar linear amplifier such as the TPA005D02 has a maximum draw of 3.25 W under the same
circumstances.
Table 3. Efficiency vs Output Power In S-V 8-0 H-Brldge Systems
Output Power (W)

Efficiency (%)

Peek Voltage (V)

0.25

63.4

2

0.145

0.5

73

2.83

0.183

0.75

77.1

3.46

0.222

1

79.3

4

0.314

4.4rt

0.3

1.25
60.6
t High peak voltages cause the THO to Increase

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Internal Dissipation (W)

TPA005D02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
class 0 effect on power supply (continued)
There is a minor power supply savings with a class 0 amplifier versus a linear amplifier when amplifying sine
waves. The difference is much larger when the amplifier is used strictly for music. This is because music has
much lower RMS output power levels, given the same peak output power (Figure 19); and although linear
devices are relatively efficient at high RMS output levels, they are very inefficient at mid-to-Iow RMS power
levels. The standard method of comparing the peak power to RMS power for a given signal is crest factor, whose
equation is shown below. The lower RMS power for a set peak power results in a higher crest factor
Crest Factor = 10 log

PPK

Prms

Time

Figure 19. Audio Signal Showing Peak and RMS Power
Figure 20 is a comparison of a 5-V class 0 amplifier to a similar linear amplifier playing music that has a 13.76-dB
crest factor. From the plot, the power supply draw from a stereo amplifier that is playing music with a 13.76 dB
crest factor is 1.02 W, while a class 0 ampljfier draws 420 mW under the same conditions. This means that just
under 2.5 times the power supply is required for a linear amplifier over a class 0 amplifier.
POWER SUPPLIED

vs
PEAK OUTPUT VOLTAGE AND PEAK OUTPUT POWER
600

SOD

i

§.

I

Co

::J

1/1

J

4D0

...,......., ~

TPA0202

3DO

200

~

~

,

--

TPAo05D02

100

o
1
0.25

1.5
0.56

........- V

2

2.5
1.56

3
2.25

/

-

~

3.5
3.06

4
4

4.5
5.06

Peak Output Voltage (V)
Peak Output Power (W)

Figure 20. Audio Signal Showing Peak and RMS Power (with Music Applied)

~ThXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-71

TPAOOSD02
2·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
class D effect on battery life
Battery operations for class D amplifiers versus linear amplifiers have similar power supply savings results. The
essential contributing factor to longer battery life is lower RMS supply current. Figure 21 compares the
TPA005D02 supply current to the supply current of the TPA0202, a 2·W linear device, while playing music at
different peak voltage levels.
SUPPLY CURRENTS

vs
PEAK OUTPUT VOLTAGE AND PEAK OUTPUT POWER
400

350
"ii'

300

g

250

E

V

C

§

TPA020i/

200

..,...., ......-V

(.)

~

""

150

:::I

..............

til

100

TPAO~5D~ ~

~

50

o
1

1.5

0.25

0.56

2
1

2.5
1.56

3
2.25

3.5
3.06

4
4

Peak Output Voltage (V)
Peak Output Power (W)

Figure 21. Supply Current vs Peak Output Voltage of TPA005D02 vs TPA0202 with Music Input
This plot shows that a linear amplifier has approximately three times more current draw at normal listening levels
than a class D amplifier, Thus, a class D amplifier has approximately three times longer battery life at normal
listening levels. If there is other circuitry in the system drawing supply current, that must also be taken into
account when estimating battery life savings.

~TEXAS

2-72

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA005D02
2·W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

---- -- -_.- .. ... ..... ..... ""' . .
A ...... LI'-'AIIVN INrvnlVl"' •• U ...
-~

crest factor and thermal considerations
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA005D02 data sheet, one can see that when the
TPA005D02 is operating from a 5-V supply into a 4-0 speaker that 4 W peaks are available. Converting Watts
to dB:
P dB = 10Log

(:w)

= 10Log

ref

(t) = 6 dB

(17)

Subtracting the crest factor restriction to obtain the average listening level without distortion yields:
6.0 dB - 18 dB
6.0 dB - 15 dB
6.0 dB - 12 dB

- 12 dB (15 dB crest factor)

= = -

9 dB (15 dB crest factor)
6 dB (12 dB crest factor)

6.0 dB - 9 dB = - 3 dB (9 dB crest factor)
6.0 dB - 6 dB = - 0 dB (6 dB crest factor)
6.0 dB - 3 dB = 3 dB (3 dB crest factor)
Converting dB back into watts:

P

W

= 10PdB/l0 x P

ref

(18)

= 63 mW (18 dB crest factor)

125 mW (15 dB crest factor)
250 mW (12 dB crest factor)
500 mW (9 dB crest factor)
= 1000 mW (6 dB crest factor)
= 2000 mW (3 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 4-0 system, the internal dissipation in the TPA005D02
and maximum ambient temperatures is shown in Table 4.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-73

TPA005D02
2·W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

crest factor and thermal considerations (continued)
Table 4. TPA005D02 Power Rating, 5-V, 4-0. StereC)
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

4

2W(3dB)

0.56

125°C

4

1000 mW (6 dB)

0.30

136°C

4

500 mW (9 dB)

0.23

139°C

4

250 mW (12 dB)

0.20

141°C

4

120 mW (15 dB)

0.14

143°C

4

63 mW (18 dB)

0.09

146°C

DISSIPATION RATING TABLE
PACKAGE

DERATING FACTOR
44.8mW/oC

5.6W

DCA

3.5W

2.9W

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
data from the dissipation rating table, the derating factor for the DCA package with 6.9 in 2 of copper area on
a multilayer PCB is 44.8 mW/oC. Converting this to 0JA:

e

JA

=

1
Derating

=

0.O~48

(19)

= 22.3°C/W

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given 0JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA005D02
is 150°C. The intemal dissipation figures are taken from the Efficiency vs Output Power graphs.
T A Max = T J Max - e JA Po

(20)

150 - 22.3(0.14 x 2)

143°C (15 dB crest factor)

150 - 22.3(0.56 x 2)

125°C (3dB crest factor)
NOTE:

Internal dissipation of 0.6 W is estimated for a 2-W system with a 15 dB crest factor per channel.
Table 4 shows that for some applications no airflow is required to keep junction temperatures in the specified
range. The TPA005D02 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Table 4 was calculated for maximum listening
volume without distortion. When the output level is reduced the numbers in the table change significantly. Also,
using 8-a speakers dramatically increases the thermal performance by increasing amplifier efficiency.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

TPA005D02
2-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS227C - AUGUST 1998 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced DCA package is based on the 56-pin TSSOP, but includes a thermal pad (see Figure 59)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Side View (a)

End View (b)

Bottom View (e)

Figure 22. Views of Thermally Enhanced DCA Package

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-75

2-76

TPAD32DD1
1D·W MONO CLASS·D AUDIO POWER AMPLIFIER
__ a

...... _."' _ _ ...

u'"'''' . . ~"'n."''"'&;.

• Extremely Efficient Class-D Mono
Operation

(TOP VIEW)

•
•
•
•
•
•

Drives Mono Speaker
10-W BTL Output Into 4 0 From 12 V
32-W Peak Music Power
Fully Specified for 12-V Operation
Low Shutdown Current
Thermally-Enhanced PowerPADTM Surface
Mount Packaging
• Thermal and Under-Voltage Protection

description
The TPA032D01 is a monolithic power IC mono
audio amplifier that operates in extremely efficient
Class-D operation, using the high switching speed
of power DMOS transistors to replicate the analog
input signal through high-frequency switching of
the output stage. This allows the TPA032D01 to
be configured as a bridge-tied load (BTL) amplifier
capable of delivering up to 10 W of continuous
average power into a 4-0 load at 0.5% THD+N
from a 12-V power supply in the high-fidelity audio
frequency range (20 Hz to 20 kHz). A BTL
configuration eliminates the need for external
coupling capacitors on the output. A chip-level
shutdown control is provided to limit total supply
current to 20 JIA., making the device ideal for
battery-powered applications.

SHUTDOWN
MUTE
AGND
INN
INP
COMP
AGND
Voo
PVoo
OUTP
OUTP
PGND
PGND
OUTN
OUTN
PVoo
Vee REG
NC
NC
AGND
PVoo
VCP
NC
CP1

10
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22

23
24

48
47
46
45
44

43
42
41
40

39
38
37
36
35
34

33
32

31
30
29
28
27
26
25

COSC
AGND
AGND
AGND
AGND
AGND
FAULTO
FAULT1
PVoo
NC
NC
PGND
PGND
NC
NC
PVoo
Vee
NC
NC
V2P5
PVoo
PGND
NC
CP2

Ne - No internal connection

The output stage is compatible with a range of power supplies from 8 V to 14 V. Protection circuitry is included
to increase device reliability: thermal and under-voltage shutdown, with a status feedback terminal for use when
,any error condition is encountered.
The high switching frequency of the TPA032D01 allows the output filter to consist of three small capacitors and
two small inductors per channel. The high switching frequency also allows for good THD+N performance.
The TPA032D01 is offered in the thermally enhanced 48-pin PowerPAD TSSOP surface-mount package
(deSignator DCA).

...
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
~ Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAO is a trademark of Texas Instruments InCOrporated.

~TEXAS .
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

Copyrtght © 2000, Texas Instruments Incorporated

2-77

'l'
ex!

-

I
I

,~ PV
INP ~

r------

PVDD

INN

~

~-~

i:i~"",
~~
ll! ~rr;I

~~~

~l"l1Gi

~~
'"

I
I
I
I,
I
I

______
----------VCP

PVDD

--

'U

~

oc

c

z~

0~

_
~

1.5 V

THERMAL
DETECT

CD

I

.:n
SHUTDOWN

PVDD

VDD

PVDD

rl>

en
cp

C
l>
c:
C

:I:

0

8'"

0

"0

:e
m
):Ii

I

5-V

I VCCREG
V2P5

RAMP
COSC ~ GENERATOR

~

~

VCP-UVLO
DETECT

~

DOUBLER
CHARGE PUMP

IJ, ____________ __________ _
J~~
~
c
c

_ __________ .1

(5j
'U

o

~

o

J

s:
"0

r-

and BIASES

NOTE B. VOO and PVOO are externally connected. AGNO and PGNO are externally connected.

(")

:::u

GATE
DRIVE

PGND

s:::W

c
s::
»
~
0

~

•

!l!

iii
m
MUTE

9~

O~
() ZO
<0
CD

PVDD
VCP

L ___

0'-

n

m
m

-

m

REGULATOR

I~~

s::
OJ

3CD

:n

CONTROL and
STARTUP
LOGIC

II

AGND

:Eo

0

~
'"
»
cI

II)
_

PVDD

COMPI

VDD~

~

c

-------------------,

GATE
DRIVE

.
•

~

0

'--1

-I'
n

::r

;;
iii
:::u

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

Terminai Functions
TERMINAL
NAME
AGND

DESCRIPTION

NO.
3,7,20,
43,44,45,
46,47

Analog ground for headphone and Class-D analog circuitry

COSC

48

CPl

24

First diode node for charge pump

CP2

25

First inverter switching node for charge pump

FAULTO

42

Logic level fauito output signal. Lower order bit of the two fault signals with open drain output.

FAULTI

41

Logic level faull1 output signal. Higher order bit of the two fault signals with open drain output.

COMP

6

Compensation capacitor terminal for Class-D amplifier

INN

4

Class-D negative input

5

Class-D positive input

INP

Connect a capacitor from analog ground to this terminal to set the frequency of the ramp reference signal.

OUTN

14,15

Class-D amplifier negative output of H-bridge

OUTP

10,11

Class-D amplifier positive output of H-bridge

PVDD

9,16

MUTE

2

Class-D amplifier power supply
Active-low TTL logic-level mute input Signal. When MUTE is held low, the selected amplifier is muted. When
MUTE is held;;> high, the device operates normally. When the Class-D amplifier is muted, the low-side output
transistors are turned on, shorting the load to ground.

18,19,23,
26,30,31,
34,35,38,
39

NC

PGND

12,13

PGND

27

Power ground for H-bridge only
Power ground for charge pump only

PGND

36,37

PVDD

21,28,33,
40
1

SHUTDOWN

Not connected

Power ground for right-channel H-bridge only.
VDD supply for charge-pump and gate drive circuitry
Active-low TTL logic-level shutdown input signal. When SHUTDOWN is held low, the device goes into shutdown
mode. When SHUTDOWN is held high, the device operates normally.

V2P5

29

2.5V internal reference bypass. This terminal requires a capacitor to ground.

VCC

32

5V supply to circuitry. This terminal is typically connected to VCCREG.

VCCREG

17

5-V regulator output. This terminal requires a l-I1F capacitor to ground for stability reasons.

VCP

22

Connect a capacitor from this terminal to power ground to provide storage for the charge pump output voltage.

VDD

8

VDD bias supply for analog circuitry. This terminal needs to be well filtered to prevent degrading the device
performance.

Class-O amplifier faults
Table 1. Class-O Amplifier Fault Table
FAULT 0

FAULT 1

1

1

No fault. The device is operating normally.

0

1

Charge pump under-voltage lock-out (VCP-UV) fault. All low-side transistors are turned on, shorting the load to
ground. Once the charge pump voltage is restored, normal operation resumes, but FAULTI is still active. This is not
a latched fault, however. FAULT1 is cleared by cycling MUTE, SHUTDOWN, or the power supply.

0

0

Thermal fault. All the low-side transistors are turned on, shorting the load to ground. Once the junction temperature
drops 20°C, normal operation resumes (not a latched fault). But the FAULTx terminals are still set and are cleared
by cycling MUTE, SHUTDOWN, or the power supply.

DESCRIPTION

~TEXAS

INSTRUMENTS
POST OFACE BOX 655303 • DALLAS. TEXAS 75265

2-79

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SL0S282A- DECEMBER 1999 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES

TSSOPi'

TA

(DCA)

t

TPA032D01 DCA
-40°C to 125°C
The DCA package Is available in left-ended tape and reel. To order
a taped and reeled part. add the suffix R to the part number (e.g .•
TPA032D01DCAR).

absolute maximum ratings over operating free-air temperature range, TC =25°C (unless otherwise
noted)t
Supply voltage, (VDD' PVoo) ............................................................... 14 V
Logic supply voltage, (Vee) ................................................................ 5.5 V
Input voltage, VI (MUTE, MODE, SHUTDOWN) ........................................ -0.3 V to 7 V
Output current, 10 (FAULTO, FAULT1), open drain terminated ................................... 1 rnA
Supply/load voltage, (FAULTO, FAULT1) ...................................................... 7 V
Charge pump voltage, Vep .......................................................... PVoo + 20 V
Continuous H-bridge output current (1 H-bridge conducting) .................................... 3.5 A
Pulsed H-Bridge output current, each output, Imax (see Note 1) .................................. 7 A
Continuous VeeREG output current, 10 (VeeREG) .......................................... 150 rnA
Continuous total power dissipation, T e 25°C ........................... See Dissipation Rating ,Table
Operating virtual junction temperature range, TJ ......•............................. -40°C to 150°C
Operating case temperature range, T e ,........................................... -40°C to 125°C
Storage temperature range, Tstg .................................................. -65°C to 260°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

=

t Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only. and
functional operation of the device at these or any other conditions beyond those indicated under "racommended operating conditions' is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: Pulse duration 10 ms, duty cycle :s: 2%

=

DISSIPATION RATING TABLE

=

=

PACKAGE

TAS25°ct
POWER RATING

DERATING FACTOR
ABOVE TA =25°C

TA 700 e
POWER RATING

TA 85°C
POWER RATING

DCA

5.6 W

44.8 mW/oC

3.6 W

2.9 W

:j: Please see the Texas Instruments document. PowerPAD Thermally Enhanced Package Application

Repol1(literature number SLMA002). for more Information on the PowerPAD package. The thermal data
was measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
MIN
Supply voltage. VDD. PVDD. LPVDD. RPVDD
Logic supply voltage. VCC
High-level input voltage. VIH (MUTE. SHUTDOWN)
Low-level input voltage. VIL (MUTE. SHUTDOWN)

NOM

V

4.5

5.5

V

2

VDD + 0.3 V

V

-0.3

0.8
1

PWM frequency

100

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 •

OAllAS. TEXAS 75265

UNIT

14

AudiO Inputs. LINN. LINP. RINN. RINP. differential input voltage

2-80

MAX

8

250

500

V
VRMS
kHZ

TPA032D01
10-W MONO CLAS5-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

electrical characteristics Class-D amplifier, VOD
See Figure 1 (unless otherwise noted)
PARAMETER

=PVDD =12 V, RL =4 (.l to 8 fl, TA =25"C,
TEST CONDmONS

Power supply rejection ratio

VDD = PVDD = XPVDD = 11 V to 13 V

IDD

Supply current

No outpu1 Iilter connected

IDD(Mute)

Supply current, mute mode

MUTE=OV

IDD(SID)

Supply current, shutdown mode

SHUTDOWN = 0 V

IIIHI

High-level input current (MUTE, MODE,
SHUTDOWN)

VIH=5.25V

IIILI

Low-level input current (MUTE, MODE,
SHUTDOWN)

VIL=-0.3V

rDS(on)

Static drain-to-source on-state resistance
(high-side + low-side FETs)

IDD=0.5A

rDS(on)

MatChing, high-side to high-side, low-side to
low-side, same channel

operating characteristics, Class-D amplifier, VOO
(unless otherwise noted)

AV

TEST CONDITIONS

Efficiency

PO= lOW,
l=lkHz

MIN

Dynamic range

18

mA

30
10

IlA
IlA

10

IlA

800

mO

98%

TYP

MAX

UNIT

W

25

dB

--eO

dB

80

dB

-50

1=1 kHz

Frequency response bandwidth, post output filter, -3 dB

Input impedance

10
20

77%

Noise floor

ZI

dB
mA

10

Gain

Maximum output power bandwidth

UNIT

=PVDD =12 V, RL =4 n, TA =25°C, See Figure 1

Output power

BOM

MAX

35

25

720

1= 1 kHz,
THD=0.5%
Device soldered on PCB,
See Note 2

Crosstalk

TYP

-40

95%

PARAMETER

Po

MIN

20

10

dB
20000

Hz

20

kHz

kQ

NOTE 2: Output power is thermally limited, TA = 23°C

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2--el

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS2B2A - DECEMBER 1999 - REVISED MARCH 2000

operating characteristics, Class-D amplifier, Voo
TA = 25°C, See Figure 2 (unless otherwise noted)
PARAMETER
Po

AV

=

PVoo

=

TEST CONDITIONS

Output power,

THO = 0.5%
Oevice soldered on PCB,
See Note 2

Efficiency

PO=7.5W,
1= 1 kHz

12 V,
MIN

RL
TYP

1= 1 kHz

BOM

Maximum output power bandwidth

ZI

Input impedance

UNIT
W

25

dB

-60

dB

80

dB

Dynamic range

,

MAX

85%

Noise Iloor

Frequency response bandwidth, post output Iilter, -3 dB

8 Q,

7.5

Gain

Crosstalk

=

-50
20

dB
20000

Hz

20

kHz
k.Q

10

NOTE 2: Output power is thermally limited, TA = 85DC

operating characteristics, Vee 5-V regulator, TA
PARAMETER

t

=25°C (unless otherwise noted)

TEST CONDITIONS

Vo

Output voltage

voo = PVoo = LPVoo = RPVoo = 8 V to 14 V,
10=Ot090mA

lOS

Short-circuit output current

Voo = PVoo = LPVoo = RPVoo = 8 V to 14 vt

MIN
4.5
90

TYP

MAX
5.5

UNIT
V
rnA

Pulse width must be limited to prevent exceeding the maximum operating virtual junction temperature 01 150DC.

thermal shutdown
PARAMETER

TEST CONDITIONS

Thermal shutdown temperature
Thermal shutdown hysteresis

~TEXAS

INSTRUMENTS
2-82

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

MIN

TYP

MAX

UNIT

165

DC

30

DC

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

PARAiviETER

iviEASUREiviEi~i iNFOR:v'AT:C~~

r---------------,

1

VeeREG

FAULTOL-R

1
-U SHUTDOWN

2 1
VeeREG ~ MUTE

FAULT1~

1
OUTN I14,1S
1

1
1
12 v!!.1§J PV
1 DO
111F
Balanced
Differential
I
I
tS '
npu Igna

1
1
11
1
1
1
V2Psn29

1

~~INN

~\:I

1

1

1
1

~

~

-=-

1000 PF T

4Q

1
INP

111F
. -_ _ _-"6'__11 eOMP

1000PF~T

111F

OUTP~1021~1-Frn~~--~~--~

1
1

{-l~

1Sl1H
---._ _- . _ - - ,

! eose

I

F
111

-

VDDt-"- 12V

1

1

1
1

1
1
VeeREG 1-1-"17'------I+-- Vee

1
1

21,28,33,34
12V

1
I PVDD
1

SOOkQ

1
ep1 1-1",,24=-----,

1

1

--4II----1_-----'3~2'__11

Tl~~

ep2 ~I=2S=--__
-'

1

To ____
Vee REG
100kQ

rO.111F

11

Vee

vep "",22=------,
1
L _______________ J
T
1--1

1

l

O.111F

Figure 1. 12-V, 4-0 Test Circuit

~TEXAS

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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-83

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r---------------,
I

FAULTOLR

I

FAULT1~

---u

I

I 14,15 30 IJ.H
OUTN i-'-'~....ryyy'-e---._-__,

VCCREG
I SHUTDOWN
VCCREG ~ MUTE

I
I

12 V9•16

I

I
I

I

PVDD

I 1-'-"0"-'-11!.-..J"YY'r-y.--_
OUTP:-

I
1 J.1F

Balanced
Differential
Input Signal

I

I

~L~l INN

V2P5!-l

.

I

~r-------"'-c!1 COMP

1000 PFT

-=- .

48

I

I

II

I

~I cosc

1000 PFT

-=7,20,43 44 45 4647
12 13273637

I

I

I
117

VCCREG t-I.!.!---1~- VCC
I
I
0.1 J.1F

AGND
PGND

r

-=- 21,28,33,34 I

I

12 V - -___- - - 1 PVDD
I

500kO

CP1 i-'12=4_---,

I

I

1
To _____...--",,32:.....1 V
VCcREG
1 CC
100kO

+~~

CP21-'125=-----1_
VCP rl2~2_-,
1

L_______________ J
Figure 2.

1~-V,

8-0 Test Circuit

~lExAs

2-84

J-

1 J,1f

VDDI-!- 12V

I
I
I

_"*----'

I

{--1~IINP
~'I
1 J.1F 6 I

1 J,1f

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

lT

0.1 J.1F

80

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SL0S282A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
VCCREG
To System {--!:--f;H~~~;----------1100 kg
Control
~ MUTE
1
1
91 1
12 V
~:!:::
:!::: ' 6 1 PVOO
10 ~F--L -;:r::- 1 ~F -;:r::- 1 ~F 1

~ V

Class-O Balanced
Oifferentlal Input
Signal

100 kg

FAULTO!---""42=----+-.....- }
1 41
To System
FAULT1
Control
1

V I I
1 ~F

1

OUTN 14,15

I
I

{ ----l~ INP
----l~

~F

INN

1
,---_ _ _....:6'--1 COMP

0,22~~ ~

4g

OUTP 1011

1
15~H
V2P51-1""29'-----l'
VOO 18
:!::: 12 V
1
11~FT

1000PF*
--L

1~0 pF T~r----'48=--: cosc

':f' ~F

r

1
,--_ _ _ _ _-'7'-2!2""0""46=47'-! AGNO
1213273637 PGNO

VCCREG

VCC

CP1 J-I!!:24=-------.
1
1 25
CP21
VCP ....,122=-----'1

l.

12 V _ _~-e--------'2"-'1-'-'!2"",8--11 PVOO
1 ~F
1
V
1
32 V
VOO
1 CC

T

=:i:=

1

L_______________ J

SOOkQ

0,22~Fh

1

47 nF

.--L-

~

0,1

~F

TO~..._ _ _•

VCCREG

100kQ

-=-

0,1~F

T
NOTE A.

~ = power ground and

-b

= analog ground

Figure 3. TPA032D01 Typical Configuration Application Circuit

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2-85

TPA032DOt
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, C,
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and ZI, the TPA032001 's input resistance forms a
high-pass filter with the corner frequency determined in equation 8.

fC(highpass) =

2~i1CI

(8)

Z, is nominally 10 kO
The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where the specification calls for a flat bass response down to 40 Hz. Equation 8 is
reconfigured as equation 9.

C I -

1

(9)

2~Zlfc

In this example, C, is 0.40 J.lF so one would likely choose a value in the range of 0.47 J.lF to 1 J.lF. A low-leakage
tantalum or ceramic capacitor is the best choice for the input capacitors. When polarized capacitors are used,
the positive side of the capaCitor should face the amplifier input, as the dc level there is held at 1.5 V, which is
likely higher than the source dc level. Please note that it is important to confirm the capacitor polarity in the
application.

differential input
The TPA032001 has differential inputs to minimize distortion at the input to the IC. Since these inputs nominally
sit at 1.5 V, dc-blocking capacitors are required on each of the four input terminals. If the signal source is
single-ended, optimal performance is achieved by treating the signal ground as a signal. In other words,
reference the signal ground at the signal source, and run a trace to the dc-blocking capacitor, which should be
located physically close to the TPA032001. If this is not feasible, it is still necessary to locally ground the unused
input terminal through a dc-blocking capacitor.

power supply decoupling, Cs
The TPAb32001 is a high-performance Class-O CMOS audio amplifier that requires adequate power supply
decoupling to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling
also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling
is achieved by using two capacitors of different types that target different types of noise on the power supply
leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-seriesresistance (ESR) ceramic capacitor, typically 0.1 J.lF placed as close as possible to the device's various Voo
leads, works best. For filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 J.lF
or greater placed near the audio power amplifier is recommended.
The TPA032001 has several different power supply terminals. This was done to isolate the noise resulting from
high-current switching from the sensitive analog circuitry inside the IC.

~TEXAS

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INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

TPA032D01
10-W MONO CLAS5-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

mute and shutdown modes
The TPA032D01 employs both a mute and a shutdown mode of operation designed to reduce supply current,
100, to the absolute minimum level during periods of non-use for battery-power conservation. The SHUTDOWN
input terminal should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN
low causes the outputs to mute and the amplifier tOEmter a low-current state, 100 = 20 IIA. Mute mode alone
reduces 100 to 10 mA.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

output filter components
The output inductors are key elements in the performance of the class-D audio amplifier system. It is important
that these inductors have a high enough current rating and a relatively constant inductance over frequency and
temperature. The current rating should be higher than the expected maximum current to avoid magnetically
saturating the inductor. When saturation occurs, the inductor loses its functionality and looks like a short circuit
to the PWM signal, which increases the harmonic distortion considerably.
A shielded inductor may be required if the class-D amplifier is placed in an EMI sensitive system; however, the
switching frequency is low for EMI considerations and should not be an issue in most systems. The dc series
resistance of the inductor should be low to minimize losses due to power dissipation in the inductor, which
reduces the efficiency of the circuit.
Capacitors are important in attenuating the switching frequency and high frequency noise, and in supplying
some of the current to the load. It is best to use capacitors with low equivalent-series-resistance (ESR). A low
ESR means that less power is dissipated in the capacitor as it shunts the high-frequency signals. Placing these
capacitors in parallel also parallels their ESR, effectively reducing the overall ESR value. The voltage rating is
also important, and, as a rule of thumb, should be 2 to 3 times the maximum rms voltage expected to allow for
high peak voltages and transient spikes. These output filter capacitors should be stable over temperature since
large currents flow through them.

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2-87

TPA032D01
10·W MONO CLASS·D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
efficiency of class·D vs linear operation
Amplifier efficiency is defined as the ratio of output power delivered to the load to power drawn from the supply.
In the efficiency equation below, PL is power across the load and Psup is the supply power.
. .

Efficiency =

'I]

PL

=--

Psup

A high-efficiency amplifier has a number of advantages over one with lower efficiency. One of these advantages
is a lower power requirement for a given output, which translates into less waste heat that must be removed
from the device, smaller power supply required, and increased battery life.
Audio power amplifier systems have traditionally used linear amplifiers, which are well known for being
inefficient. Class-D amplifiers were developed as a means to increase the efficiency of audio power amplifier
systems.
A linear amplifier is designed to act as a variable resistor network between the power supply and the load. The
transistors operate in their linear region and voltage that is dropped across the transistors (in their role as
variable resistors) is lost as heat, particularly in the output transistors.
The output transistors of a class-D amplifier switch from full OFF to full ON (saturated) and then back again,
spending very little time in the linear region in between. As a result, very little power is lost to heat because the
transistors are not operated in their linear region. If the transistors have a low on-resistance, little voltage is
dropped across them, further reducing losses. The ideal class-D amplifier is 100% efficient, which assumes that
both the on-resistance (rDS(on)) and the switching times of the output transistors are zero.
the ideal class-D amplifier

To illustrate how the output transistors of a class-D amplifier operate, a half-bridge application is examined first
(see Figure 4).
VDD

J

M1

~

I~
+

L

J

M2

Rl
clI

VOUT

cT

-=Figure 4_ Half-Bridge Class-D Output Stage

Figures 5 and 6 show the currents and voltages of the half-bridge circuit. When transistor M1 is on and M2 is
off, the inductor current is approximately equal to the supply current. When M2 switches on and M1 switches
off, the supply current drops to zero, but the inductor keeps the inductor current from dropping. The additional
inductor current is flowing through M2 from ground. This means that VA (the voltage at the drain of M2, as shown
in Figure 4) transitions between the supply voltage and slightly below ground. The inductor and capacitor form
a low-pass filter, which makes the output current equal to the average of the inductor current. The low pass filter
averages VA, which makes VOUT equal to the supply voltage multiplied by the duty cycle.

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INSTRUMENTS
POST OFFICE BOX 655303 e, DALLAS. TEXAS 75265

TPA032D01
10·W MONO CLASS·D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

APpliCATiON iNFORiviATiON
the ideal class-D amplifier (continued)
Control logic is used to adjust the output power, and both transistors are never on at the same time. If the output
voltage is rising, M1 is on for a longer period of time than M2.
Inductor Current
t--:::.~+-"""",:--::""~""""",~-:::;;;o~+-""",,,,-...,""-t---""..::---::;;o~-

Output Current

Supply Current

I
(J

O~-~----~-~--~--4----~-~--~---'

M1 onl M1 offl M1 onl
M2 offl M2 on I M2 offl • • •
Time

Figure 5. Class-D Currents

Voo

'----"

...--.. I"---- V- '---

- --

t("VA

~You T

o
M1 on IM1 off IM1 onl
M20fflM20n IM20ffl···
TIme

Figure 6. Class-D Voltages

~TEXAS

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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-£9

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS2B2A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
the ideal clas$-D amplifier (continued)
Given these plots, the efficiency of the class-D device can be calculated and compared to an ideal linear
amplifier device. In the derivation below, a sine wave of peak voltage (Vp) is the output from an ideal class-D
and linear amplifier and the efficiency is calculated.

LINEAR

CLASS-D
Vp

VL(rms)

Vp

= 12

VL(rms)

A
I ) = IL(rms)Vx VL(rms)
verage (00
00

P _
L -

V

= 12
L(rms)
RL

2

V 2

= _P_
2 RL

2 Vp
Average (100 ) = 1t x R
L
Psup = Voo x Average(loo)
P

-

SUP -

Efficiency

Efficiency

Voox IL(rms) x VL(rms)
Voo

------';c;-'-----'--..!-

=

=

Tj

Tj

PL

=Psup

=1

Psup

= Voo

x Average ( 100)

=

Voo Vp
2
R
x 1t
L

PL
Efficiency = Tj = - Psup

Efficiency = Tj

Efficiency

V

= Tj = ~4 x ~
V
DD

In the ideal efficiency equations, assume that Vp = Voo, which is the maximum sine wave magnitude without
clipping. Then, the highest efficiency that a linear amplifier can have without clipping is 78.5%. A class-D
amplifier, however, can ideally have an efficiency of 100% at all power levels.
The derivation above applies to an H-bridge as well as a half-bridge. An H-bridge requires approximately twice
the supply current but only requires half the supply voltage to achieve the same output power-factors that
cancel in the efficiency calculation. The H-bridge circuit is shown in Figure 7.

voo

J

voo

L

+ VOUT-

l

Rl

J

-::-

Figure 7. H-Bridge Class-D Output Stage

~TEXAS

2-90

L

Tel
-::-

INSTRUMENTS
POST OFACE BOX 655303 • DALLAS. TEXAS 75265

TPA032D01
10·W MONO CLASS·D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

ApPliCATiON iNFORiviATiON
losses in a real·world class-O amplifier
Losses make class-O amplifiers nonideal, and reduce the efficiency below 100%. These losses are due to the
output transistors having a nonzero r08(on), and rise and fall times that are greater than zero.
The loss due to a nonzero r08(on) is called conduction loss, and is the power lost in the output transistors at
nonswitching times, when the transistor is on (saturated). Any r08(on) above 0 n causes conduction loss.
Figure 8 shows an H-bridge output circuit simplified for conduction loss analysis and can be used to determine
new efficiencies with conduction losses included.
VOO=12V

rOS(on)

0.36 r.!

5 Mr.!

rOS(off)

0.36 r.!

rDS(on)

RL
4r.!
rOS(off)

5 MO

Figure 8. Output Transistor Simplification for Conduction Loss Calculation
The power supplied, P8UPo is determined to be the power output to the load plus the power lost in the transistors,
assuming that there are always two transistors on.
PL
Efficiency = 11 = - P8UP
Efficiency

= 11
12 2r 08(on)

Efficiency = 11

+ 12RL

RL
2r 08(on)

+ RL

n, RL = 4 n)
= 85% (at all output levels r 08(on) = 0.36 n, RL = 4 n)

Efficiency = 11 = 95% (at all output levels r 08(on) = Q.1
Efficiency = 11

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TPA032D01
1Q-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS282A- DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
losses In a real-world class-D amplifier (continued)

Losses due to rise and fall times are called switching losses. A diagram of the output, showing switching losses,
is shown in Figure 9.

H

tswon

+

H

tswoff

=

tsw

Figure 9. Output Switching Losses

Rise and fall times are greater than zero for several reasons. One is that the output transistors cannot switch
instantaneously because (assuming a MOSFET) the channel from drain to source requires a specific period
of time to form. Another is that transistor gate-source capacitance and parasitic resistance in traces form RC
time constants that also increase rise and fall times.
Switching losses are constant at all output power levels, which means that switching losses can be ignored at
high power levels in most cases. At low power levels, however, switching losses must be taken into account
when calculating efficiency. Switching losses are dominated by conduction losses at the high output powers,
but should be considered at low powers. The switching losses are automatically taken into account if you
consider the quiescent current with the output filter and load.
class-D effect on power supply

Efficiency calculations are an important factor for proper power supply design in amplifier. systems. Table 2
shows Class-D efficiency at a range of output power levels (per channel) with a 1-kHz sine wave input. The
maximum power supply draw from a stereo 10-W per channel audio system with 4-0 loads and a 12-V supply
is almost 26 W. A similar linear amplifier such as the TPA032D01 has a maximum draw of greater than 50 W
under the same circumstances.
Table 2. Efficiency vs Output Power in 12;-V 4-0 H-Bridge Systems
Output Power (W)

Efficiency (%)

Peak Voltage (V)

Internal Dissipation (W)

.0.5
2

41.7

2
4

0.35

66.7
75.1
5
78
8
10
77.9
t High peak voltages cause the THO to increase

6.32
8
6.94t

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0.5
0.83
1.13
1.42

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SL0S282A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
class-D effect on power supply (continued)

There is a minor power supply savings with a class-O amplifier versus a linear amplifier when amplifying sine
waves. The difference is much larger when the amplifier is used strictly for music. This is because music has
much lower RMS output power levels, given the same peak output power (see Figure 10); and although linear
devices are relatively efficient at high RMS output levels, they are very inefficient at mid-to-Iow RMS power
levels. The standard method of comparing the peak power to RMS power for a given signal is crest factor, whose
equation is shown below. The lower RMS power for a set peak power results in a higher crest factor
Crest Factor = 10 log

PPK

Prm.

TIme

Figure 10. Audio Signal Showing Peak and RMS Power

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2-93

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

crest factor and thermal considerations
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA032D01 data sheet, one can see that when the
TPA032D01 is operating from a 12-V supply into a 4-0 speaker that 20-W peaks are available. Converting watts
to dB:
PdB = 10Log

(:w)

= 10Log

ref

(~O) = 6 dB

(17)

Subtracting the crest factor restriction to obtain the average listening level without distortion yields:
6.0 dB - 18 dB
6.0 dB - 15 dB
6.0 dB - 12 dB

- 12 dB (15 dB crest factor)

= = -

9 dB (15 dB crest factor)
6 dB (12 dB crest factor)

6.0 dB - 9 dB

- 3 dB (9 dB crest factor)

6.0 dB - 6 dB

= - 0 dB (6 dB crest factor)
= 3 dB (3 dB crest factor)

6.0 dB - 3 dB

Converting dB back into watts:

Pw

=

1OPdBj10 x P

ref

(18)

= 315 mW (18 dB crest factor)

=

630 mW (15 dB crest factor)

= 1.25 W (12 dB crest factor)

= 2.5 W (9 dB crest factor)
= 5 W (6 dB crest factor)
= 10 W (3 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 10 W of continuous power output with a 3 dB
crest factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 12-V, 4-0 system, the internal dissipation in the
TPA032D01 and maximum ambient temperatures are shown in Table 3.

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TPA032D01
10·W MONO CLASS·D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

APPliCATiON iNFORiviATION

crest factor and thermal considerations (continued)
Table 3. TPA032D01 Power Rating, 12-V, 4-0, Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

20

10W(3dB)

2.84

87°C

20

5W(6dB)

1.66

113°C
125°C

20

2.5W(9dB)

1.12

20

1.25 W (12 dB)

0.87

125°C

20

630 mW (15 dB)

0.7

125°C

20

315 mW (18 dB)

0.6

125°C

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
data from the dissipation rating table, the derating factor for the DCA package with 6.9 in 2 of copper area on
a multilayer PCB is 44.8 mWrC. Converting this to ElJA:

1
9 JA

(19)

Derating

=_1_
0.0448

=

22.3°C/W

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given ElJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA032D01
is 150°C. The internal dissipation figures are taken from the Efficiency vs Output Power graphs.
TA Max = T J Max - 9 JA P D

(20)

150 - 22.3(0.35)

125°C (15 dB crest factor)

150 - 22.3(1.42)

118°C (3dB crest factor)

(Maximum recommended case temperature is 125°C)
NOTE:
Intemal dissipation of 0.7 W is estimated for a 1O-W system with a 15 dB crest factor per channel.

The TPA032D01 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. Table 3 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-0
speakers dramatically increases the thermal performance by increasing amplifier efficiency.

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2-95

TPA032D01
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS282A - DECEMBER 1999 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced DCA package is based on the 56-pin TSSOP, but includes a thermal pad (see Figure 11)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Thermal
Pad

Side View (a)

End View (b)

I f

Bottom View (e)

Figure 11. Views of Thermally Enhanced DCA Package

~TEXAS

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POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA032D02
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
- REVISED MARCH 2000

UCA I'ACKAGE

• Extremely Efficient Class-D Stereo
Operation

(TOP VIEW)

•
•
•
•
•
•

Drives Land R Channels
10-W BTL Output Into 4 0 From 12 V
32-W Peak Music Power
Fully Specified for 12-V Operation
Low Shutdown Current
Thermally-Enhanced PowerPADTM Surface
Mount Packaging
• Thermal and Under-Voltage Protection

SHUTDOWN
MUTE
AGND
LINN
LlNP
LCOMP
AGND

description
The TPA032D02 is a monolithic power IC stereo
audio amplifier that operates in extremely efficient
Class-D operation, using the high switching speed
of power DMOS transistors to replicate the analog
input signal through high-frequency switching of
the output stage. This allows the TPA032D02 to
be configured as a bridge-tied load (BTL) amplifier
capable of delivering up to 10 W of continuous
average power into a 4-0 load at 0.5% THD+N
from a 12-V power supply in the high-fidelity audio
frequency range (20 Hz to 20 kHz). A BTL
configuration eliminates the need for external
coupling capacitors on the output. A Chip-level
shutdown control is provided to limit total supply
current to 20 1lA, making the device ideal for
battery-powered applications.

Voo
LPVoo
LOUTP
LOUTP
PGND
PGND
LOUTN
LOUTN
LPVoo
Vee REG
NC
NC
AGND
PVoo
VCP
NC
CP1

10
2
3
4
5
6
7
8

9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24

48

47
46
45
44
43

42
41
40

39
38
37
36
35
34
33

32
31
30
29
28
27
26
25

COSC
AGND
AGND
RINN
RINP
RCOMP
FAULTO
FAULT1
RPVoo
ROUTP
ROUTP
PGND
PGND
ROUTN
ROUTN
RPVoo
Vee
NC
NC
V2P5
PVoo
PGND
NC
CP2

Ne - No internal connection

The output stage is compatible with a range of power supplies from 8 V to 14 V. Protection circuitry is included
to increase device reliability: thermal and under-voltage shutdown, with a status feedback terminal for use when
any error condition is encountered.
The high switching frequency of the TPA032D02 allows the output filter to consist of three small capaCitors and
two small inductors per channel. The high switching frequency also allows for good THD+N performance.
The TPA032D02 is offered in the thermally enhanced 48-pin PowerPAD TSSOP surface-mount package
(designator DCA).

A.

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

2-97

U>

~

r--------------------------

I
I

~

g

C

CD

a5
------------------.

LPVOO

VCP

~~ I~~

~

c

3
!!l.

n

LlNP~

IfI

10kO

I

~

....
£~z~
~-

~
ll!

d
l§t::~

~~~
I

OETECT

MUTE

-=-

GATE
ORIVE
5-V

I
I

1----iIVeeREG

I Vee

RPVOO
VCP

RAMP

PVOO

-=-

-=-

GENERATOR

I
I

VCp·UVLO

GATE
ORIVE

OETECT

RCOMP
RINP .,!.t----i.

.1-/

RINNII

I
IfokO

II
RPVoO IL-

I

OkO
1

OOUBLER
CHARGE PUMP

PVOO

RPVoo

GATE
ORIVE

________ _____ ---------i3
~
<

PGNO

I
I
I
I
I
I
I
I
I
I
I
I

I

1.5V

Q
1...---

I
AGNOn

RPVOO
VCP

:.

__

~
~g

i3

NOTE B. LPVOO. RPVOO. and PVOO are externally connected. AGND and PGNO are externally connected.

~

__

~

~

__

~

~

I
I
I
I
________ JI

~

"'jl!
~o
(1)(0)

~N
mS
::rJN
m
0
(')

~
C

~

c:

52

§ 0-a

PVOO

REGULATOR
and BIASES

I
COSC 1

;;:
:D

I

I

0

0
::J:

PVOD

VOor- VOO

:S
U>

LPVOO
•

I
I

I
:D

m

m

LOGIC

VCP

LCOMPI

SHUTDOWN

CONTROL and
STARTUP

10kO

I

~

i

THERMAL

GATE
ORIVE

1.5V

0

[!l

!II:D

PVOO

LPVOO, LPVOO

LlNN~
I

I

.... ~

V2P5

~
::rJ

I:-a
r-

:Ii
m
::rJ

TPA032D02
1Q.W STEREO CLASS-D AUDIO POWER AMPLIFIER
Sl0S243A - DECEMBER 1999 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME
AGND

DESCRIPTION

NO.

3,7,20,
46,47

Analog ground for Class·D analog circuitry

COSC

48

Connect a capacitor from analog ground to this terminal to set the frequency of the ramp reference signal.

CPl

24

First diode node for charge pump

CP2

25

First inverter switching node for charge pump

FAULTO

42

Logic level faullO output signal. Lower order bit of the two fault signals with open drain output.

FAULTl

41

Logic level faultl output signal. Higher order bit of the two fault signals with open drain output.

LCOMP

6

Compensation capacitor terminal for left-channel Class-D amplifier

LINN

4

Class-O left-channel negative input

LlNP

5

Class-O left-channel positive input

LOUTN

14,15

Class-O amplifier left-channel negative output of H-bridge

LOUTP

10, 11

Class-O amplifier ieft-channel positive output of H-bridge

LPVOO

9, 16

Class-O amplifier left-channel power supply

MUTE

2

Active-low TTL logic-level mute input signal. When MUTE is held low, the salected amplifier is muted. When MUTE
is held> high, the device operates normally. When the Class-O amplifier is muted, the low-side output transistors
are tumed on, shorting the load to ground.

NC

18,19,
23,26,
30,31

No connection

PGNO

12,13

Power ground for left-channel H-bridge only

PGND

27

PGNO

36,37

Power ground for right-channel H-bridge only

PVOO

21,28

VOO supply for charge-pump and gate drive circuitry

Power ground for charge pump only

RCOMP

43

Compensation capacitor terminal for right-channel Class-D amplifier

RINN

45

Class-O right-channel negative input

RINP

44

RPVOO

33,40

Class-O right-channel positive input
Class-O amplifier right-channel power supply

ROUTN

34,35

Class-O amplifier right-channel negative output of H-bridge

ROUTP

38,39

Class-O amplifier right-channel positive output of H-bridge

SHUTDOWN

1

VCC

32

5V supply to logic. This terminal is typically connected to VCCREG.

VCCREG

17

5-V regulator output. This terminal requires a l-I1F capacitor to ground for stability reasons.

V2P5

29

2.5V internal reference bypass. This terminal requires a capaCitor to ground.

VCP

22

Connect a capacitor from this terminal to power ground to provide storage for the charge pump output voltage.

VOO

8

VOD bias supply for analog circuitry. This terminal needs to be well filtered to prevent degrading the device
performance.

Active-low TTL logic-level shutdown input signal. When SHUTDOWN is held low, the device goes into shutdown
mode. When SHUTDOWN is held high, the device operates normally.

~TEXAS

\

\

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

2-99

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

Class-D amplifier faults
Table 1. Class-O Amplifier Fault Table
FAULT 0

FAULT 1

1

1

No fault. The device is operating normally.

0

1

Charge pump under-voltage lock-out (VCP-UV) fault. All low-side transistors are tumed on, shorting the load to
ground. Once the charge pump voltage is restored, normal operation resumes, but FAULT1 is still active. This is not
a latched fault, however. FAULT1 is cleared by cycling MUTE, SH)JTDOWN, or the power supply.

0

0

Thermal fault. All the low-side transistors are tumed on, shorting the load to ground. Once the junction temperature
drops 20°C. normal operation resumes (not a latched fault). But the FAULTx terminals are still set and are cleared
by cycling MUTE, SHUTDOWN, or the power supply.

DESCRIPTION

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

TSSOJ>t
(DCA)

-40°C to 125°C

TPA032D02DCA

t The DCA package is available in left-ended tape and reel. To order
a taped and reeled part, add the suffix R to the part number (e.g.,
TPA032D02DCAR).

~TEXAS

INSTRUMENTS
2--100

POST OFFICE BOX 655303 • DAL.l,.AS, TEXAS 75265

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A- DECEMBER 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range, TC = 25 C (uniess oinerwi::.e
noted)t
v

Supply voltage, (Voo, PVoo, LPVoo, RPVoo) ...............................................• 14 V
Logie supply voltage, (Vce> ................................................................ 5.5 V
Input voltage, VI (MUTE, MODE, SHUTDOWN) ........................................ -0.3 V to 7 V
Output current, 10 (FAULTO, FAULT1), open drain terminated ................................... 1 mA
Supply/load voltage, (FAULTO, FAULT1) ...................................................... 7 V
Charge pump voltage, Vcp .......................................................... PVoo + 20 V
Continuous H-bridge output current (1 H-bridge conducting) .................................... 3.5 A
Pulsed H-Bridge output current, each output, Imax (see Note 1) .................................. 7 A
Continuous VccREG output current, 10 (VcCREG) .......................................... 150 mA
Continuous total power dissipation, T C 25°C ........................... See Dissipation Rating Table
Operating virtual junction temperature range, TJ .................................... -40°C to 150°C
Operating case temperature range, T C ............................................ -40°C to 125°C
Storage temperature range, Tstg .................................................. -65°C to 260°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

=

t

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: Pulse duration = 10 ms, duty cycle s 2%
DISSIPATION RATING TABLE
PACKAGE

TA:;;25°C*
POWER RATING

DERATING FACTOR
ABOVE TA 25°C

TA = 70°C
POWER RATING

TA =85°C
POWER RATING

DCA

5.6W

44.8mW/OC

3.6W

2.9W

=

:I: Please

see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application
Report (literature number SLMA002), for more information on the PowerPAD package. The thermal data
was measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
MIN
Supply voltage, VDD, PVDD, LPVDD, RPVDD
Logic supply voltage, VCC
High-level input voltage, VIH (MUTE, SHUTDOWN)
Low-level input voltage, VIL (MUTE, SHUTDOWN)

NOM

UNIT

14

V

4.5

5.5

V

2

VDD + 0.3 V

V

-0.3

0.8

Audio inputs, LINN, LlNP, RINN, RINP, differential input voltage
PWM frequency

MAX

8

1
100

250

500

V
VRMS
kHZ

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-101

TPA032D02
10-W STEREO CLASSoD AUDIO POWER AMPLIFIER.
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

=

electrical characteristics Class-D amplifier, Voo PVoo
TA = 25°C, See Figure 1 (unless otherwise noted)
PARAMETER

=LPVoo =RPVoo =12 V, RL =4 a to 8 a,

TEST CONDmoNS

Power supply rejection retio

VOO = PVOO = xPVOO = 11 V to 13 V

100

Supply current

No output filter connected

IOO(Mute)

Supply current, mute mode

MUTE"OV

IOO(SIDI

Supply current, shutdown mode

SHUTDOWN = 0 V

IIIHI

High-level input current (MUTE, MODE,
SHUTDOWN)

VIH=5.25V

IIILI

Low-level input current (MUTE, MODE,
SHUTDOWN)

VIL=-0.3V

rDS(on)

Static drein-to-source on-state resistance
(high-side + low-side FETs)

IDO=0.5A

rDS(on)

Matching, high-side to high-side, low-side to
low-side, same channel

operating characteristics, Class-D amplifier, Voo
TA = 25"C, See Figure 1 (unless otherwise noted)

AV

TEST CONDITIONS

Efficiency

PO=10W,
f= 1 kHz

MIN

92%

Noise floor

f=1kHz

Frequency response bandwidth, post output filter, -3 dB

rnA

10

IIA
IIA

10

IIA

800

mO

98%

TYP

MAX

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75Z65

UNIT

W

dB

95%

-eo

dB

80

dB
dB

-50
20

10

NOTE 2: Output power is thermally limited, TA = 23°C

2-102

18

30

77%

Dynamic range

Input impedance

10
20

25

Left/right channel gain matching

ZI

dB

rnA

10

Gain

Crosstalk

UNIT

=PVoo =LPVoo =RPVoo =12 V, RL =4 a,

Output power

Maximum output power bandwidth

MAX
35

25

720

f=1 kHz,
THO;' 0.5%, per channel,
Device soldered on PCB,
See Note 2

BOM

TYP

-40

95%

PARAMETER

Po

MIN

20000

Hz

20

kHz

kO

TPA032D02
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

operating characteristics, Class·D amplifier, Voo = PVoo = LPVoo = RPVoo = 12 V, RL = 8
TA 25°C, See Figure 2 (unless otherwise noted)

=

PARAMETER

Po

AV

TEST CONDITIONS

Output power,

THD = 0.5%, per channel,
Device soldered on PCB,
See Note 2

Efficiency

PO=7.5W,
f= 1 kHz

MIN

TYP

7.5

W

25
92%

Noise floor
f= 1 kHz

Frequency response bandwidth, post output filter, -3 dB

-eo

dB

80

dB
dB

-50
20

Maximum output power bandwidth
Input impedance

20000

Hz

20

kHz
kQ

10

ZI
NOTE 2: Output power is thermally limited, TA = 85°C

operating characteristics, Vee 5-V regulator, TA
PARAMETER

dB

95%

Dynamic range
Crosstalk

UNIT

85%

Gain
LeIVright channel gain matching

BOM

MAX

n,

=25°C (unless otherwise noted)

TEST CONDITIONS

Vo

Output voltage

VDD = PVDD = LPVDD = RPVDD = 8 V to 14 V,
10=Oto90mA

lOS

Short-circuit output current

VDD = PVDD = LPVDD = RPVDD = 8 V to 14 vt

MIN

4.5
90

TYP

MAX

5.5

UNIT

V
rnA

t Pulse width must be limited to prevent exceeding the maximum operating virtual junction temperature of 150°C.

thermal shutdown
TEST CONDITIONS

PARAMETER

Thermal shutdown temperature
Thermal shutdown hysteresis

MIN

TYP

MAX

UNIT

165

°C

30

°C

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-103

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r---------------,

1

FAULTO~

1

FAULT1~

1

1

-.!.J

VCCREG
1 SHUTDOWN
VCC REG ~ MUTE

1

1 1415 15 J!H
LOUTN ~2.!'"'--''YYY"'I....,..-.-~...-----,

1
1

12 V 9,16
1 J!F
Balanced
Differential
Input Signal

I

LPVDD

1
1

{-1~1

t--!i

LlNP

-1
LINN
1 J!F
..-_ _ _-=.6--11 LCOMP

r l1

I

43

1000pF-L

J

1000 PF T
-=-

I

RCOMP

1

rjcosc

I

1000 PF T

1 J!F

;-=:_-'+

RINN

1

I

33,34 RPVDD
3,7,20,46,47 1 AGND
12,13,27,36,37 PGND

l

i

21 28

T

1
1PVDD

1
1
1
1
To VCCREG _ _ _ _.--3=2'-! V
1 cc
500kn

100kn

1

ROUTP

1L _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ ~1

3839

Figure 1, 12-V, 4-0 Test Circuit

-!I1TEXAS

2-104

0.1 J!F

1

-1~

12 V

12V

ToVCC

1
{-1~1 RINP
1 J!F

f-

T

1

-=-

Balanced
Differential
Input Signal

1

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

47 nF

0.1J!F

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r---------------,

I

FAULTO~

I

FAULT1~

I
V REG
ee
Vee REG

~
--LJ

I

SHUTDOWN
MUTE

I
30l1H
LOUTN 11'-'1'-='415
,='---fY'CY'"--*_ _ _ _ _ _- - - ,

I

I
I

12 V 9,16

I

111F

I
I

I
I

I

LPVDD

LOUTP 1-1
-,-,,0,,-,1.....
1 --"''YY~_ _' ' ' ' ' ' ' _ - - '

I

Balanced {-1t--L!1 LINP
Differential
Input Signal
-1~1 LINN
111F
______~6~1
[I LeOMP

~
-Li

1000 pF --L

T-=-

1000PFT

II
In9
V2P5

ReOMP

~
-=-

111F

Balanced
Differential
Input Signal

VDD~

~\I
1 I1F

12 V

33,34

3,7,20,46,47
12,13,27,36,37

1

12V

To
VeeREG

r

I

I

0.1 I1F

I

RINP

e p1 1 24
I

RINN

-..LT 47nF

CP2 (-'I2=5'----__
-'

I
I RPVDD
I AGND

I
I
I
I
I

l

vep 1-'12=2,-------.,

PGND

-

I
I
_ _--._--=2'-"1'-=2=--8I PVDD
I
I
500kO
I
I
_ _--.--tf-----=3"'--2I V
I ee
100kO

12V

VeeREG 1-11.!.!7----4I--- To Vee

I

i

11

I
I
I

I!
I
I

~~

1 F

I-

I

{-1~

J

I

-=I
~~se

1000 pF T

I
I

T

0.111F

I

ROUTN

I
I
I
I
I

3435

80

ROUTP l-3=8""3,,,-9--f'rTY~_ _- - ' _ - - '

I

IL _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ I
~

Figure 2. 12-V, 8-n Test Circuit

-!11 TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-105

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 -' REVISED MARCH 2000

APPLICATION INFORMATION
To System {--~-r;H~~~;;;;----------l

~ MUTE

Control

12 V

.......
10J.lF J....

~

1
VCCREG
1
1
1
1 100kn

1
1
:::t: :::t:9,16 1 LPVDD
~ 1 J.lF ~ 1 J.IF 1
1
1

{--1t---Ll
1 J.lF

Left Class-D Balanced
Differential Input
Signal

Control
15 J.lH

1'1 LINN

1
1 LCOMP

1 J.lF

J-.-

6

rl

1000pF-r

-=

To System
LINP

4 I

----.J f

4'1

RCOMP

1
1

1000PFr

-=

I

l00kn

FAULTO;-I4.:.::2'----+____- }

1COSC

48

1
1
1 J.lF
1
{--1~ RINP

1000PFr

-=

Right Class-D Balanced
Differential Input
Signal

----.J f 45 I

1'1 RINN
1 J.lF

12 V

.......

:::t:

£3,34
10 F J.... -;::c- 1 J.lF -;::c- 1 J.lF

J.I~

V

V
37204647
1213273637
21,28

1
1 RPVDD
1
1
AGND

1 PGND
11PVDD
t-=_----'+47nF

l

~-----.--I 0,111F
0,1 J.lF

32

VDD

1 J.IF

Vee

500kn
To ---t.....- - - .
VCCREG
l00k'1

T

0,1 J.lF

NOTE A.

.& =power ground and -b

=analog ground

Figure 3. TPA032D02 Typical Configuration Application Circuit

~TEXAS

2-106

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

4'1

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, C,
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, C, and Z" the TPA032002's input resistance forms a
high-pass filter with the corner frequency determined in equation 8.

fC(highpass) =

2:rt~ICI

(8)

Z, is nominally 10 k.Q

The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where the specification calls for a flat bass response down to 40 Hz. Equation 8 is
reconfigured as equation 9.

CI

=

_1_

(9)

2:rtZ l f c

In this example, C, is 0.40 !iF so one would likely choose a value in the range of 0.47 j.LF to 1 !iF. A low-leakage
tantalum or ceramic capacitor is the best choice for the input capacitors. When polarized capacitors are used,
the positive side of the capacitor should face the amplifier input, as the dc level there is held at 1 .5 V, which is
likely higher than the source dc level. Please note that it is important to confirm the capacitor polarity in the
application.

differential input
The TPA032002 has differential inputs to minimize distortion at the input to the IC. Since these inputs nominally
sit at 1.5 V, dc-blocking capaCitors are required on each of the four input terminals. If the signal source is
single-ended, optimal performance is achieved by treating the signal ground as a signal. In other words,
reference the Signal ground at the signal source, and run a trace to the dc-blocking capacitor, which should be
located physically close to the TPA032002. If this is not feasible, it is still necessary to locally ground the unused
input terminal through a dc-blocking capacitor.

power supply decoupling, Cs
The TPA032002 is a high-performance Class-O CMOS audio amplifier that requires adequate power supply
decoupling to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling
also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling
is achieved by using two capacitors of different types that target different types of noise on the power supply
leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-seriesresistance (ESR) ceramic capacitor, typically 0.1 j.LF placed as close as possible to the device's various Voo
leads, works best. For filtering lower-frequency noise Signals, a larger aluminum electrolytiC capacitor of 10 !iF
or greater placed near the audio power amplifier is recommended.
The TPA032002 has several different power supply terminals. This was done to isolate the noise resulting from
high-current switching from the sensitive analog Circuitry inside the IC.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

2-107

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

mute and shutdown modes
The TPA032D02 employs both a mute and a shutdown mode of operation designed to reduce supply current,
100, to the absolute minimum level during periods of non-use for battery-power conservation. The SHUTDOWN
input terminal should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN
low causes the outputs to mute and the amplifier to enter a low-current state, 100 = 20 j.LA. Mute mode alone
reduces 100 to 10 mAo

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

output filter components
The output inductors are key elements in the performance of the class-D audio amplifier system. It is important
that these inductors have a high enough current rating and a relatively constant inductance over frequency and
temperature. The current rating should be higher than the expected maximum current to avoid magnetically
saturating the inductor. When saturation occurs, the inductor loses its functionality and looks like a short circuit
to the PWM signal, which increases the harmonic distortion considerably.
A shielded inductor may be required if the class-D amplifier is placed in an EMI sensitive system; however, the
switching frequency is low for EMI considerations and should not be an issue in most systems. The dc series
resistance of the inductor should be low to minimize losses due to power dissipation in the inductor, which
reduces the efficiency of the circuit.
Capacitors are important in attenuating the switching frequency and high frequency noise, and in supplying
some of the current to the load. It is best to use capacitors with low equivalent-series-resistance (ESR). A low
ESR means that less power is dissipated in the capacitor as it shunts the high-frequency signals. Placing these
capacitors in parallel also parallels their ESR, effectively reducing the overall ESR value. The voltage rating is
also important, and, as a rule of thumb, should be 2 to 3 times the maximum rms voltage expected to allow for
high peak voltages and transient spikes. These output filter capacitors should be stable over temperature since
large currents flow through them.

~TEXAS

2-108

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

efficiency of class-O vs linear operation
Amplifier efficiency is defined as the ratio of output power delivered to the load to power drawn from the supply.
In the efficiency equation below, PL is power across the load and Psup is the supply power.
Efficiency = 11

P

= __L_
P sup

A high-efficiency amplifier has a number of advantages over one with lower efficiency. One of these advantages
is a lower power requirement for a given output, which translates into less waste heat that must be removed
from the device, smaller power supply required, and increased battery life.
Audio power amplifier systems have traditionally used linear amplifiers, which are well known for being
inefficient. Class-O amplifiers were developed as a means to increase the efficiency of audio power amplifier
systems.
A linear amplifier is designed to act as a variable resistor network between the power supply and the load. The
transistors operate in their linear region and voltage that is dropped across the transistors (in their role as
variable resistors) is lost as heat, particularly in the output transistors.
The output transistors of a class-O amplifier switch from full OFF to full ON (saturated) and then back again,
spending very little time in the linear region in between. As a result, very little power is lost to heat because the
transistors are not operated in their linear region. If the transistors have a low on-resistance, little voltage is
dropped across them, further reducing losses. The ideal class-O amplifier is 100% efficient, which assumes that
both the on-resistance (rDS(On) and the switching times of the output transistors are zero.

the ideal class-D amplifier
To illustrate how the output transistors of a class-O amplifier operate, a half-bridge application is examined first .
(see Figure 4).

Voo

J

M1

~

I~
+

L

J

M2

RL

clI cr

vOUT

Figure 4. Half-Bridge Class-D Output Stage
Figures 5 and 6 show the currents and voltages of the half-bridge circuit. When transistor M1 is on and M2 is
off, the inductor current is approximately equal to the supply current. When M2 switches on and M1 switches
off, the supply current drops to zero, but the inductor keeps the inductor current from dropping. The additional
inductor current is flowing through M2 from ground. This means that VA (the voltage atthe drain of M2, as shown
in Figure 4) transitions between the supply voltage and slightly below ground. The inductor and capacitor form
a low-pass filter, which makes the output current equal to the average of the inductor current. The low pass filter
'
averages VA, which makes VOUT equal to the supply voltage multiplied by the duty cycle.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-109

TPA032D02
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
the ideal class-O amplifier (continued)

Control logic is used to adjust the output power, and both transistors are never on at the same time. If the output
voltage is rising, M1 is on for a longer period of time than M2.

I--::.o~p......."--:;o~+"""",-o--:~q.:!!oo,,,c-~~P~--::;o~-

Output Current

Supply Current

....~--~--....~--~....~----....--------.

o~---+

M1 onl M1 offl M1 onl
M2 offl M2 on I M2 offl • • •
Time

Figure 5. Class-O Currents

~""~--~""-r---,~""r---~""-r---'''''---VDD

VOUT

....~---L........~--~....~--~

O~---+""",,~--r-

M1 on IM1 off IM1 onl
M20ff lM20n 1M2 off I • • •
Time

Figure 6. Class-O Voltages

~TEXAS

2-110

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
the ideal class-D amplifier (continued)
Given these plots, the efficiency of the class-O device can be calculated and compared to an ideal linear
amplifier device. In the derivation below, a sine wave of peak voltage (Vp) is the output from an ideal class-O
and linear amplifier and the efficiency is calculated.

CLASS-D

LINEAR

Vp
V L(rms) =

Vp

.f2

VL(rms) =

A
I ) = 'L(rms)Vx VL(rms)
verage (DD
DD

P

.f2

_ VL(rms)
L RL

2

V p2
2 RL

2 Vp
Average ('DD ) = it x R

L

P sup = V DD x Average ( 'DD ) =

P sup = V DD x Average(, DD )

P sup =

VDDX 'L(rms) x VL(rms)
V DD

PL
Efficiency = 11 = - P sup

Efficiency

=

11

=1

VDD Vp
2
R
x it
L

PL
Efficiency = 11 = - P sup

Efficiency

V

Efficiency = 11 =!! x -E...
4 VOO

In the ideal efficiency equations, assume that Vp = VDD, which is the maximum sine wave magnitude without
clipping. Then, the highest efficiency that a linear amplifier can have without clipping is 78.5%. A class-O
amplifier, however, can ideally have an efficiency of 100% at all power levels.
The derivation above applies to an H-bridge as well as a half-bridge. An H-bridge requires approximately twice
the supply current but only requires half the supply voltage to achieve the same output power-factors that
cancel in the efficiency calculation. The H-bridge circuit is shown in Figure 7.

Voo

Voo
+ VOUT-

L
L

Figure 7. H-Bridge Class-D Output Stage

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

2-111

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

losses In a real-world class-O amplifier
Losses make class-D amplifiers nonideal, and reduce the efficiency below 100%. These losses are due to the
output transistors having a nonzero rOS(on), and rise and fall times that are greater than zero.
The loss due to a nonzero rOS(on) is called conduction loss, and is the power lost in the output transistors at
nonswitching times, when the transistor is on (saturated). Any rOS(on) above 0 n causes conduction loss.
Figure 8 shows an H-bridge output circuit simplified for conduction loss analysis and can be used to determine
new efficiencies with conduction losses included.
VOO=12V

1
rOS{on)

0.360

5MO

rOS{off)

0.360

rOS{on)

RL
40
rOS{off)

5MO

I
Figure 8. Output Transistor Simplification for Conduction Loss Calculation
The power supplied, PsuP, is determined to be the power output to the load plus the power lost in the transistors,
assuming that there are always two transistors on.
Efficiency

=

TJ --

Efficiency

=

TJ

Efficiency = TJ
Efficiency = TJ
Efficiency = TJ

PL

PsuP

12 2rOS(on)

+ 12RL

RL
2rOS(on)

+ RL

n, RL = 4 n)
= 85% (at all output levels r OS(on) = 0.36 n, RL = 4 n)
= 95%

(at all output levels rOS(on)

= 0.1

~TEXAS

2-112

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

TPA032D02
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
losses in a real-world class-D amplifier (continued)
Losses due to rise and fall times are called switching losses. A diagram of the output, showing switching losses,
is shown in Figure 9.

H

tswon

+

tSWoff

=

tsw

Figure 9. Output Switching Losses
Rise and fall times are greater than zero for several reasons. One is that the output transistors cannot switch
instantaneously because (assuming a MOSFET) the channel from drain to source requires a specific period
of time to form. Another is that transistor gate-source capacitance and parasitic resistance in traces form RC
time constants that also increase rise and fall times.
Switching losses are constant at all output power levels, which means that switching losses can be ignored at
high power levels in most cases. At low power levels, however, switching losses must be taken into account
when calculating efficiency. Switching losses are dominated by conduction losses at the high output powers,
but should be considered at low powers. The switching losses are automatically taken into account if you
consider the quiescent current with the output filter and load.

class-D effect on power supply
Efficiency calculations are an important factor for proper power supply design in amplifier systems. Table 2
shows Class-D efficiency at a range of output power levels (per channel) with a 1-kHz sine wave input. The
maximum power supply draw from a stereo 10-W per channel audio system with 4-0 loads and a 12-V supply
is almost 26 W. A similar linear amplifier such as the TPA032D02 has a maximum draw of greater than 50 W
under the same circumstances.

Table 2. Efficiency vs Output Power in 12-V 4-0 H-Sridge Systems
Output Power (W)

Efficiency (%)

Peak Voltage (V)

Internal Dissipation (W)

0.5

41.7

2

0.7

2

66.7

4

1.0

5

75.1

6.32

1.66

8

78

8
8.94t

2.26

10
77.9
t High peak voltages cause the THO to Increase

2.84

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

2-113

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
class-D effect on power supply (continued)
There is a minor power supply savings with a class-O amplifier versus a linear amplifier when amplifying sine
waves. The difference is much larger when the amplifier is used strictly for music. This is because music has
much lower RMS output power levels, given the same peak output power (see Figure 10); and although linear
devices are relatively efficient at high RMS output levels, they are very inefficient at mid-to-Iow RMS power
levels. The standard method of comparing the peak power to RMS power for a given signal is crest factor, whose
equation is shown below. The lower RMS power for a set peak power results in a higher crest factor
Crest Factor

=

10 log

PPK

P nns

Time

Figure 10. Audio Signal Showing Peak and RMS Power

~TEXAS

INSTRUMENTS
2-114

POST OFFICE BOX 655303 • OALLAS. TEXAS 75265

TPA032D02
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA032D02 data sheet, one can see that when the
TPA032D02 is operating from a 12-V supply into a 4-Q speaker that 20-W peaks are available. Converting watts
to dB:
P dB = 10Log

(:w) =

10Log

ref

(~O) = 6 dB

(17)

Subtracting the crest factor restriction to obtain the average listening level without distortion yields:
6.0 dB - 18 dB

- 12 dB (15 dB crest factor)

6.0 dB - 15 dB = - 9 dB (15 dB crest factor)
6.0 dB - 12 dB

= - 6 dB (12 dB crest factor)

6.0 dB - 9 dB = - 3 dB (9 dB crest factor)
6.0 dB - 6 dB

= -

6.0 dB - 3 dB

= 3 dB (3 dB crest factor)

0 dB (6 dB crest factor)

Converting dB back into watts:

Pw

= 1OPdB/10 x

P

ref

(18)

= 315 mW (18 dB crest factor)

=

630 mW (15 dB crest factor)

= 1.25 W (12 dB crest factor)

=

2.5 W (9 dB crest factor)

= 5 W (6 dB crest factor)

=

10 W (3 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the .
amplifier system. Comparing the absolute worst case, which is 10 W of continuous power output with a 3 dB
crest factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 12-V, 4-Q system, the internal dissipation in the
TPA032D02 and maximum ambient temperatures are shown in Table 3.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

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TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

crest factor and thermal considerations (continued)
Table 3. TPA032D02 Power Rating, 12-V, 4-0, Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

20

10W(3dB)

2.84

23°C

20

5W(6dB)

1.66

75°C

20

2.5W(9dB)

1.12

100°C

20

1.25 W (12 dB)

0.87

111°C

20

630 mW (15 dB)

0.7

118°C

20

315 mW (18 dB)

0.6

123°C

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
data from the dissipation rating table, the derating factor for the DCA package with 6.9 in 2 of copper area on
a multilayer PCB is 44.8 mW/oC. Converting this to ElJA:

1
Derating

(19)

=_1_
0.0448
= 22.3°C/W

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given ElJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA032D02
is 150°C. The internal dissipation figures are taken from the Efficiency vs Output Power graphs.
TA Max

T J Max - 9 JA Po

(20)

150 - 22.3(0.7 x 2) = 118°C (15 dB crest factor)
150 - 22.3(2.84 x 2)

= 23°C (3dB crest factor)
NOTE:

Internal dissipation of 1.4 W is estimated for a 10-W system with a 15 dB crest factor per channel.

The TPA032D02 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. Table 3 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-n
speakers dramatically increases the thermal performance by increasing amplifier efficiency.

~TEXAS

2-116

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

TPA032D02
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS243A - DECEMBER 1999 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced DCA package is based on the 56-pin TSSOP, but includes a thermal pad (see Figure 11)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

(c=x

I

DIE

f1 tj E5 Ej E5 d

Side View (a)

End View (b)

Bottom View (c)

Figure 11. Views of Thermally Enhanced DCA Package

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-117

2-118

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
- DECEMBER 1999 - REVISED MARCH 2000

• Extremely Efficient Class-D Mono
Operation
• Drives Mono Speaker, Plus Stereo
Headphones
• 10-W BTL Output Into 4 0 From 12 V
• 32-W Peak Music Power
• Fully Specified for 12-V Operation
• Low Shutdown Current
• Class-AB Headphone Amplifier
• Thermally-Enhanced PowerPADTM Surface
Mount Packaging
• Thermal and Under-Voltage Protection
description

DCA PACKAGE
(TOP VIEW)

SHUTDOWN
MUTE
MODE
INN
INP
COMP
AGND
VDD
PVDD
OUTP
OUTP
PGND
PGND
OUTN
OUTN
PVDD
HPREG
HPLOUT
HPLIN
AGND
PVDD
VCP
HPDL
CP1

10
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16

48
47
46
45
44

43
42
41
40
39
38
37

36
35
34

COSC
AGND
AGND
AGND
AGND
AGND
FAULTO
FAULT1
PVOD
NC
NC
PGND
PGND
NC
NC
PVOD
HPVcc
HPROUT
HPRIN
V2P5
PVOD
PGND
HPDR
CP2

The TPA032D03 is a monolithic power IC mono
33
audio amplifier that operates in extremely efficient
17
32
Class-D operation, using the high switching speed
18
31
of power DMOS transistors to replicate the analog
19
30
input signal through high-frequency switching of
20
29
the output stage. This allows the TPA032D03 to
21
28
be configured as a bridge-tied load (BTL) amplifier
22
27
capable of delivering up to 10 W of continuous
23
26
average power into a 4-0 load at 0.5% THD+N
24
25
from a 12-V power supply in the high-fidelity audio
frequency range (20 Hz to 20 kHz). A BTL
NC - No internal connection
configuration eliminates the need for external
coupling capacitors on the output. Included is a Class-AB headphone amplifier with interface logic to select
between the two modes of operation. Only one amplifier is active at any given time, and the other is in
power-saving sleep mode. Also, a chip-level shutdown control is provided to limit total supply current to 20 ~,
making the device ideal for battery-powered applications.
The output stage is compatible with a range of power supplies from 8 V to 14 V. Protection circuitry is included
to increase device reliability: thermal and under-voltage shutdown, with a status feedback terminal for use when
any error condition is encountered.
The high switching frequency of the TPA032D03 allows the output filter to consist of three small capacitors and
two small inductors per channel. The high switching frequency also allows for good THD+N performance.
The TPA032D03 is offered in 'the thermally enhanced 48-pin PowerPAD TSSOP surface-mount package
(designator DCA) .

.A.
.a..

Please be aware that an important notice concerning availability. standard warranty. and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

Copyright © 2000, Texas Instruments Incorporated

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-119

~

I\)

o

_____
r------II

p

g
-------.

YDD,- U'Vno

------.

... f.------I

VCP

INN~
I
:

If

~~
iVl

Z

VDD~

3

----,I

I
I
I
I

PVoo
THERMAL
DETECT

GATE
DRIVE

VDD

I

HPVCC

•

______ _
--

--------

I HPVCC

c

HP
DEPOP

C§

"
NOTE B. VOO and PVOD are externally connected. AGND and PGNO are externally connected.

HPROUT

g

c

~

OJ

m

(')

CD
CD
CD

J:-o
en

:D

0

S;

r-

cp
C

J:-o

c:

c

::J:

0

0

:3

0

"'1:r

:e
m
J:-o

s:::

r-

DOUBLER
CHARGE PUMP

I:

ZO

;;:

"'1:r

HPLOUT

L.--PGND

O~

OW

~

I HPREG

,.--------11 HPUN

~

:eo
s:::W

m

0

~

_______

.........

cr~

:;;
iii
~

VCp..UVLO
DETECT

I ..1
AGND L-

0m

;;:

-=-

Ul

o-

MUTE

V2PS
RAMP
GENERATOR

~

0I

MODE

and BIASES

COSC

llil

DI
_

I

-=-

~V
REGULATOR

CJ)

m
<
Cii
m

PVDD
PVDD

0

:D

f =SH""U""TDO=W=N

CONTROL and
STARTUP
LOGIC

•

I
I

CD

PVDD

1.SV
VCP

~.

j

()

GATE
DRIVE

COMPI

o_~
:!lz
fa (/)
~~~
~t:~m
':l~

!:i

10kn! 10kn

I

@

PVDD

~

or

it!
c:

it!

c:

0

o:::r

'HPRIN

I HPDL

0-----"
-- J
...

HPDR

TPA032DOS
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME
AGND

DESCRIPTION

NO.
7,20,
43,44,
45,46,47

Analog ground for headphone and Class-D analog sections

CaMP

6

Compensation capacitor terminal for Class-D amplifier

COSC

48

Connect a capacitor from analog ground to this terminal to set the frequency of the ramp reference signal.

CPl

24

First diode node for charge pump

CP2

25

First inverter switching node for charge pump

FAULTO

42

Logic level faultO output signal. Lower order bit of the two fauH signals with open drain output.

FAULTI

41

Logic level fault1 output signal. Higher order bit of the two fault signals with open drain output.

HPDL

23

Depop control for left headphone

HPDR

26

Depop control for right headphone

HPLIN

19

Headphone amplifier left input

HPLOUT

18

Headphone amplifier left output

HPREG

17

5-V regulator output. This terminal requires a 1-IlF capacitor to ground for stability reasons.

HPRIN

30

Headphone amplifier right input

HPROUT

31

Headphone amplifier right output

HPVCC

32

5V supply to headphone amplifier and logic. This terminal is typically connected to HPREG.

INN

4

INP

5

Class-D positive input

MODE

3

TTL logic-level mode input signal. When MODE is held low, the main Class-D amplifier is active. When MODE is
held> high, the head phone amplifier is active.

MUTE

2

Active-low TTL logic-level mute input signal. When MUTE is held low, the selected amplifier is muted. When MUTE
is held> high, the device operates normally. When the Class-D amplifier is muted, the low-side output transistors
are turned on, shorting the load to ground.

NC

34,35,

Class-D negative input

No connection

38,39
OUTN

14,15

Class-D amplifier negative output of H-bridge

OUTP

10,11

Class-D amplifier positive output of H-bridge

PGND

12,13

PGND

27

PGND

36,37

PVDD

9,16,21,
28,33,40

SHUTDOWN

1

Power ground for H-bridge only
Power ground for charge pump only
Power ground for H-bridge only
VDD supply for charge-pump, headphone regulator, Class-D amplifier, and gate drive Circuitry
Active-low TTL logic-level shutdown input signal. When SHUTDOWN is held low, the device goes into shutdown
mode. When SHUTDOWN is held high, the device operates normally.

V2P5

29

VCP

22

Connect a capacitor from this terminal to power ground to provide storage for the charge pump output voHage.

VDD

8

VDD bias supply for analog circuitry. This terminal needs to be well fiHered to prevent degrading the device
performance.

2.5V internal reference bypass. This terminal requires a capacitor to ground.

~·TEXAS

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2-121

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

Class-D amplifier faults
Table 1. Class-D Amplifier Fault Table
FAULT 0

FAULT 1

1

1

No fault. The device is operating normally.

0

1

Charge pump under·voltage lock-out (VCP-UV) fault. All low-side transistors are turned on, shorting the load to
ground. Once the charge pump voltage is restored, normal operation resumes, but FAULT1 is still active. This is not
a latched fault, however. FAULT1 is cleared by cycling MUTE, SHUTDOWN, or the power supply.

0

0

Thermal fault. All the low-side transistors are turned on, shorting the load to ground_ Once the junction temperature
drops 20°C, normal operation resumes (not a latched fault). But the FAULTx terminals are still set and are cleared
by cycling MUTE, SHUTDOWN, or the power supply.

DESCRIPTION

headphone amplifier faults
The thermal fault remains active when the device is in head phone mode. This fault operation has exactly the
same as it does for the Class-D amplifier (see Table 1).
If HPVCC drops below approximately 4.5 V, the head phone is disabled. Once HPVcc exceeds approximately
4.5 V, the head phone amplifier is re-enabled. No fault is reported to the user.
AVAILABLE OPTIONS
PACKAGED DEVICES
TA

TSSOP"t
(DCA)

-40°C to 125°C

TPA032D03DCA

t The DCA package is available in left-ended tape and reel. To order
a taped and reeled part, add the suffix R to the part number (e.g.,
TPA032D03DCAR).

~TEXAS

2-122

INSTRUMENTS
POST OFFICE BOX 655303 • OALLAS, TEXAS 75265

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

=

absolute maximum ratings over operating free-air temperature range, TC 25°C (uniess utherwise
noted)t
Supply voltage, (Voo, PVoo) ............................................................... 14 V
Headphone supply voltage, (HPVcc) ........................................................ 5.5 V
Input voltage, VI (MUTE, MODE, SHUTDOWN) ........................................ -0.3 V to 7 V
Output current, 10 (FAULTO, FAULT1), open drain terminated ................................... 1 mA
Supply/load voltage, (FAULTO, FAULT1) ...................................................... 7 V
Charge pump voltage, Vcp .......................................................... PVoo + 20 V
Continuous H-bridge output current (1 H-bridge conducting) .................................... 3.5 A
Pulsed H-Bridge output current, each output, Imax (see Note 1) .................................. 7 A
Continuous HPREG output current, 10 (HPREG) ............................................ 150 mA
Continuous total power dissipation, T C = 25°C ........................... See Dissipation Rating Table
Operating virtual junction temperature range, TJ .................................... -40°C to 150°C
Operating case temperature range, Tc ............................................ -40°C to 125°C
Storage temperature range, Tstg .................................................. -65°C to 260°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: Pulse duration = 10 ms. duty cycle ,,; 2%
DISSIPATION RATING TABLE
PACKAGE

TA:;;25°C*
POWER RATING

DCA

5.6W

DERATING FACTOR
ABOVE TA = 25°C

TA = 70°C
POWER RATING

TA = 85°C
POWER RATING

3.6W

2.9W

:I: Please

see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application
Report (literature number SLMA002), for more information on the PowerPAD package. The thermal data
was measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
MIN
Supply voltage, VDD, PVDD
Headphone supply voltage, HPVCC
High-level input voltage, VIH (MUTE. MODE, SHUTDOWN)
Low-level input voltage, VIL (MUTE, MODE, SHUTDOWN)

NOM

UNIT

14

V

4.5

5.5

V

2

VDD + 0.3 V

V

-0.3

0.8

Audio inputs, LINN, LlNP. RINN, RINP, HPLIN. HPRIN. differential input voltage
PWM frequency

MAX

8

1
100

250

500

V
VRMS
kHZ

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-123

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

electrical characteristics Class-D amplifier, Voo
See Figure 1 (unless otherwise noted)
PARAMETER

=PVoo =12 V, RL =4 Q to 8 Q, TA =25°C,
TEST CONOITIONS

Power supply rejection ratio

VOO = PVOO= 11 Vto 13V

100

Supply current

No output filter connected

IOO(Mute)

Supply current, mute mode

MUTE=OV

IOOISID)

Supply current, shutdown mode

SHUTDOWN = 0 V

IIIHI

High-level input current (MUTE, MOOE,
SHUTDOWN)

VIH=5.25V

IIILI

Low-level input current (MUTE, MOOE,
SHUTDOWN)

VIL=-0.3V

rOS(on)

Static drain-to-source on-state resistance
(high-side + low-side FETs)

IOO=0.5A

rOS(on)

Matching, high-side to high-side, low-side to
low-side, same channel

operating characteristics, Class-D amplifier, Voo
(unless otherwise noted)
PARAMETER

Po

AV

MIN

25

95%

TEST CONDmONS

Efficiency

PO=10W,
1=1 kHz

MIN

Dynamic range
1= 1 kHz

Frequency response bandwidth, post output filter, -3 dB

Input impedance

10

18

mA

20

30
10

IJA
IJA

10

IJA

800

mO

98%

TYP

-!!1
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

MAX

UNIT

W

25

dB

-60

dB

80

dB

-50
20

10

NOTE 2: Output power is thennally limited, TA = 23°C

2-124

mA

77%

Noise Iloor

ZL

dB
35

10

Gain

Maximum output power bandwidth

UNIT

=PVoo =12 V, RL =4 Q, TA =25°C, See Figure 1

Output power

BOM

MAX

-40

720

f= 1 kHz,
THO = 0.5%,
Oevice soldered on PCB,
See Note 2

Crosstalk

TYP

dB
20000

Hz

20

kHz

kO

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

operating characteristics, Ciass-D ampiifier, YDD = FYDD = 12 'Y,"L
(unless otherwise noted)
PARAMETER
Po

AV

=0 Q, TA =26"C, See ~igi.ii'e 2

TEST CONDITIONS

Output power,

THD = 0.5%
Device soldered on PCB,
See Note 2

Efficiency

PO=7.5W,
f= 1 kHz

MIN

Dynamic range
f= 1 kHz

Z,

Input impe,dance

W

25

dB

-60

dB

80

dB

-50

Frequency response bandwidth, post output filter, -3 dB
Maximum output power bandwidth

UNIT

85%

Gain

BOM

MAX

7.5

Noise floor

Crosstalk

TVP

20

dB
20000

Hz

20

kHz
kg

10

NOTE 2: Output power is thermally limited, TA = 85°C

electrical characteristics, headphone amplifier, HPVCC
(unless otherwise noted)

=5 V, RL = 32 n, TA = 25°C, See Figure 3

PARAMETER

TEST CONDmONS

MIN

Power supply rejection ratio

TVP

MAX

UNIT

-10

VN

-60
-1

Uncompensated gain range

dB

'DD

Supply current .

9

12

mA

IDQ(MUTE)

Supply current, mute mode

9

12

rnA

'DDISlDl

Supply current, shutdown mode

20

30

jJA

operating characteristics, headphone amplifier, HPVCC
TA = 25°C, See Figure 3 (unless otherwise noted)
PARAMETER
Po

= 5V,

RL

=32 n, gain set at -10VN,

TEST CONDITIONS

Output power

THD=0.5%,
f= 1 kHz

Crosstalk

f = 1 kHz

Frequency response bandwidth, post output filter, -3 dB
BOM

Maximum output power bandwidth

ZI

Input impedance

MIN

TVP

MAX

50

mW
dB

-60
20

UNIT

20

kHz

20

kHz
Mg

>1

operating characteristics, HPREG S-V regulator, TA = 25°C (unless otherwise noted)
PARAMETER

t

TEST CONDITIONS

MIN

Vo

Output voltage

VDD = PVDD = LPVDD = RPVDD = 8 V to 14 V,
'0=Ot090mA

4.5

lOS

Short-circuit output current

VDD = PVDD = LPVDD = RPVDD = 8 V to 14 vt

90

TVP

MAX
5.5

UNIT
V
rnA

Pulse width must be limited to prevent exceeding the maximum operating virtual junction temperature of 150°C.

thermal shutdown
PARAMETER

TEST CONDITIONS

Thermal shutdown temperature
Thermal shutdown hysteresis

MIN

TVP

MAX

UNIT

165

°C

30

°C

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

2-125

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION
r---------------~

FAULTO~

I

1

~I SHUTDOWN
HPREG
HPREG ~ MUTE

FAUU1~
I

I 1415 1511H
OUTN ;.-.:-:"-,,=--"'YYyY'-_~,-----,

~MODE
I
I
I
I
I

111F
Balanced
DIfferential
Input Signal

{--1~
~~

i \ i INN

I

111F

L

61 eOMP

1000PFT

1

-=

rjeose

1000PFT

1

-=

f

INP

1
rO.

I

7 20 46 47 I AGND
12,13:27:36:37 1 PGND

-

9,16,21,28,33,34
12 V - - - - - - - - - - - / PVDD

i.~I

5001Ul

30

t-=---'+

HPLIN

-L

II
I

L _______________ J

I

-=
Figure 1. 12-V, 4-0 Test Circuit

~TEXAS

2-126

47 nF

vep!~

HPRIN

To
- 32 I
HPREG - -____-+---==--i HPVee
100kQ

1I1F

-=

I
I
I

~

To HPVee

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

T

-=

O,111F

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

PARAiviETER MEASUREiviENT iNFOiiiviAT;CN
r---------------~

FAULTO~

I

I

I

I

-..!J

HPREG
I SHUTOOWN
HPREG ~ MUTE

.r4
-=
I
11lF

Balanced
Differential
Input Signal

MODE

I

I
I

I
I
I

I
I

OUTpI1011
I

{---1~

INP

---1~

INN

-=

I
!l9
V2P5 1
I
11lF

I

I

r:
48

I cosc

I

I
1000PFT
I
I
-=
I
7,20,46,47 I AGND
12,13,27,36, 37 PGNO

1
-=

12V

J

HPLOUT~

H;-

HPROUT
HPREG '-11,,-,-7--~I"'-- To HPVCC
I
HPDR~
0.11lF

1
I

T

I

I

-= 30
r
!I

8

VOD I--"'-- 12 V
I

123

HPDL~

9,16,21,28,33,34 I
I PVOD
~ HPLIN
500kQ

8Q

1

11lF 6 1
.1r-------=--:1 COMP
1000PFT

FAULT1~
II 1415 30llH
OUTN ;-:-:"-"-=--fY'{"Y'-..-__..-_--,

I
CP11 24

l

T

I
CP21-12",5'--__
-'

HPRIN

47nF

-= 32 I
Vcp;-I2=2,-----,
To -----+---+--=:.....11 HPVCC
I
I
HPREG
I
I
....L
I
I
L
_
_
_
_
_
_
_
_
_
_
_
_
_
_
_
J
T
0.11l F
100kQ

Figure 2. 12-V, 8-0 Test Circuit

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-127

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r---------------,

-!...J

HPREG

SHUTDOWN

-L.I

HPREG
HPREG

i

MUTE
3 1 MODE

r-=~

FAULT1 1
OUTN 1 14,15

1

..r..---"-;I INP
-=-

1 42

1

1

5

FAULTO

OUTP

~INN

1

J--!ChlL

1

r-I

I1

. -_ _-=---"6:-;1 COMP

L

V2P5 29

r - - - l l

1000 pF T

.--e48
=-;

:70PF~T

COSC

I

--L.

1

T

1 j1F

18
-=VDDr- 12V

.-_--'7-"',2""0,""48"",4:!.!.7-1 AGND
1._-,1=2,,-,-,13=.2,,-,7.=36=.3,",-7; PGND

1 18
32j1F
HPLOUT t-I..:..:....--'-----1
HPROUT 1 31
~~-,

I

HPREG

J-!Z+ HPVCC

1

HPDR t-=26~_ _--,
1
2 3'--_ _ _ _ _ _-'
HPDL J-=

9,16,21.28.33.34IpVDD

12V

HPLOUT

1

I

l

1001<01
1
~ f-V\II.,........o---_1!-"9'-! HPLIN
100 1<0
1
0.1j1F

Left SE
HP Input
RlghtSE
HPlnput

100kO

CP11 24
1
CP2 J-I=25'--__
--'
1
VCP t-I=22'-_1""

T

1

1
1

1

I

~ f-V\II.,........o----,3",,0,-! HPRIN

0.1j1F 100
VDD
1<0
5001<0

To

32

HPV

1

1L _ _cc
_ _ _ _ _ _ _ _ _ _ _ _ _ J1

HPROUT

HPREG ---41.....- - -..
100kO
-=-

0.1j1F

T
Figure 3. Headphone Test Circuit

2-128

-!I
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

....L
T

47nF

0.1j1F

320

TPA032D03
10-W MONO CLASS-D AUDIO POWER AMPLIFIER
SLOS283A - DECEMBER 1999 - REVISED MARCH 2000

. --- .- ---_ ........................" . .

A ...... LI",." IV .... U'IrvnIVI"'IIVI'I

HPREG
To ~~~~~:.

{-+i;H~;;;;;----------'100 kg
l00kQ
MUTE
I 42
.--.vvv~___--=3-11 MODE
FAULTOII -=--+----4~} ToSystem
~
~ ::!::
::!::9,16 II PVDD
FAULT11----'-''--____
t---Control

4

100 kQ

12V

10!iF

t

"7 1 !iF"71 !iF 1
1 !iF

Left Class-D Balanced
Differential Input
Signal

1
OUTN 1-'--14:..<..,l:..::S--.J"YT'\ryt--_--<~-_,

I

{ -1~

INP

1

-1~

INN
1 !iF
1
,--_ _ _--"6'-11 COMP

1000PFf

-=-

I

1000 pF

48

T

-L

-=-

lO high, the head phone amplifier is active.

MUTE

2

Active-low TTL logic-level mute input signal. When MUTE is held low, the selected amplifier is muted. When MUTE
is held> high, the device operates normally. When the Class-D amplifier is muted, the low-side output transistors
are tumed on, shorting the load to ground.

PGND

12,13

PGND

27

PGND

36,37

Power ground for right-channel H-bridge only

PVDD
RCOMP

21,28

VDD supply for charge-pump, headphone regulator, and gate drive circuitry

43

Class-D amplifier left-channel power supply

Power ground for left-channel H-bridge only
Power ground for charge pump only

Compensation capacitor terminal for right-channel Class-D amplifier

RINN

45

Class-D right-channel negative input

RINP

44

Class-D right-channel positive input

RPVDD

33,40

Class-D amplifier right-channel power supply

ROUTN

34,35

Class-D amplifier right-channel negative output of H-bridge

ROUTP

38,39

Class-D amplifier right-channel positive output of H-bridge

SHUTDOWN

1

Active-low TTL logic-level shutdown input signal. When SHUTDOWN is held low, the device goes into shutdown
mode. When SHUTDOWN is held high, the device operates normally.

V2P5

29

2.5V internal reference bypass. This terminal requires a capacitor to ground.

VCP

22

Connect a capacitor from this terminal to power ground to provide storage for the charge pump output voltage.

VDD

8

VDD bias supply for analog circuitry. This terminal needs to be well filtered to prevent degrading the device
performance.

-!!1
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-143

TPA032D04
10-W STEREO CLAS5-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

Class-D amplifier faults
Table 1. Class-D Amplifier Fault Table
FAULT 0

FAULT 1

1

1

No fault. The device is operating normally.

DESCRIPTION

0

1

Charge pump under-voHage lock-out (VCP-UV) fauH. All low-side transistors are turned on, shorting the load to
ground. Once the charge pump voltage is restored, normal operation resumes, but FAULT1 is still active. This is not
a latched fauH, however. FAULT1 is cleared by cycling MUTE, SHUTDOWN, or the power supply.

0

0

Thermal fault. All the low-side transistors are turned on, shorting the load to ground. Once the Junction temperature
drops 20°C, normal operation resumes (not a latched fault). But the FAULTx terminals are still set and are cleared
by cycling MUTE, SHUTDOWN, or Ihe power supply.

headphone amplifier faults
The thermal fault remains active when the device is in head phone mode. This fault operation has exactly the
same as it does for the Class-O amplifier (see Table 1).
If HPVCC drops below approximately 4.5 V, the head phone is disabled. Once HPVcc exceeds approximately
4.5 V, the head phone amplifier is re-enabled. No fault is reported to the user.
AVAILABLE OPTIONS
PACKAGED DEVICES
TA

TSSOP"t
(DCA)

-40°C 10 125°C

TPA032D04DCA

t The DCA package Is available In left-ended lape and reel. To order
a taped and reeled part, add the suffix R to the part number (e.g.,
TPA032D04DCAR).

~1ExAs

2-144

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range, TC =25°C (unless otherwise
noted)t
Supply voltage, (Voo, PVoo, LPVoo, RPVoo) ............................................... 14 V
Headphone supply voltage, (HPVcc) ........................................................ 5.5 V
Input voltage, VI (MUTE, MODE, SHUTDOWN) ........................................ -0.3 V to 7 V
Output current, 10 (FAULTO, FAULT1), open drain terminated ................................... 1 mA
Supplylload voltage, (FAULTO, FAULT1) ...................................................... 7 V
Charge pump voltage, Vcp .......................................................... PVoo + 20 V
Continuous H-bridge output current (1 H-bridge conducting) .................................... 3.5 A
Pulsed H-Bridge output current, each output, Imax (see Note 1) .................................. 7 A
Continuous HPREG output current, 10 (HPREG) ............................................ 150 mA
Continuous total power dissipation, T C 25°C ........................... See Dissipation Rating Table
Operating virtual junction temperature range, TJ .................................... -40°C to 150°C
Operating case temperature range, T C ............................................ -40°C to 125°C
Storage temperature range, Tstg .................................................. -65°C to 260°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

=

t

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: Pulse duration = 10 ms, duty cycle s 2%
DISSIPATION RATING TABLE

=1=

PACKAGE

TA S 25°ct
POWER RATING

DERATING FACTOR
ABOVE TA 25°C

TA = 70°C
POWER RATING

TA = 85°C
POWER RATING

DCA

5.6W

44.8mW/oC

3.6W

2.9W

=

Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application
Report (literature number SLMA002), for more information on the PowerPAD package. The thermal data
was measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
MIN
Supply voltage, VDD, PVDD, LPVDD, RPVDD
Headphone supply voltage, HPVCC
High-level input voltage, VIH (MUTE, MODE, SHUTDOWN)
Low-level input voltage, VIL (MUTE, MODE, SHUTDOWN)

NOM

UNIT

14

V

4.5

5.5

V

2

VDD+0.3V

V

-0.3

0.8

Audio inputs, LINN, LlNP, RINN, RINP, HPLlN, HPRIN, differential input voltage
PWM frequency

MAX

8

1
100

250

500

V
VRMS
kHZ

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 752~5

2-145

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S203A - DECEMBER 1999 - REVISED MARCH 2000

=

electrical characteristics Class-D amplifier, Voo PVoo
TA = 25°C, See Figure 1 (unless otherwise noted)

=LPVoo =RPVoo =12 V, RL =4 g to 8 il,

TEST CONDITIONS

PARAMETER

Power supply rejection ratio

VDD = PVDD = xPVDD = 11 V to 13 V

IDD

Supply current

No output filter connected

MIN

TYP

MAX

-40
25

UNIT

dB
35

rnA

IDD(Mute)

Supply current, mute mode

MUTE=OV

10

18

rnA

IDD(SID)

Supply current, shutdown mode

SHUTDOWN = 0 V

20

30

IIIHI

High-level input current (MUTE, MODE,
SHUTDOWN)

VIH=5.25V

10

ItA
ItA

Illll

low-level input current (MUTE, MODE,
SHUTDOWN)

Vll=-0.3V

10

ItA

rDS(on)

Static drain-to-source on-state resistance
(high-side + low-side FETs)

IDD=0.5A

800

mil

rDS(on)

Matching, high-side to high-side, low-side to
low-side, same channel

operating characteristics, Class-D amplifier, Voo
TA = 25°C, See Figure 1 (unless otherwise noted)

95%

AV

TEST CONDITIONS

Output power

f=1kHz,
THD = 0.5%, per channel,
Device soldered on PCB,
See Note 2

Efficiency

PO= 10W,
f=1kHz

MIN

92%

Dynamic range
Crosstalk

Input impedance

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • OALlAS, TEXAS 75265

dB

95%
-60

dB

80

dB

20

10

NOTE 2: Output power is thermally lim~ed, TA = 23°C

2-146

W

-60

f= 1 kHz

Frequency response bandwidth, post output filter, -3 dB

UNIT

77%

Noise floor

ZI

MAX

25

lefVright channel gain matching

Maximum output power bandwidth

TYP

10

Gain

BOM

98%

=PVoo =LPVoo =RPVoo =12 V, RL =4 il,

PARAMETER

Po

720

dB
20000

Hz

20

kHz

kn

TPA032D04
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

operating characteristics, Class-O amplifier, VDD
TA = 25°C, See Figure 2 (unless otherwise noted)

=PVDD =LPVDD =RPVDD =12 V, RL =8 U,

PARAMETER
Po

TEST CONDITIONS

Output power,

THD = 0.5%, per channel,
Device soldered on PCB,
See Note 2

Efficiency

PO=7.5W,
1= 1 kHz

MIN

MAX

7.5

W

25

Leftlright channel gain matching

92%

Noise floor
Dynamic range
Crosstalk

1=1 kHz

BOM

Maximum output power bandwidth

ZI

Input impedance

dB

95%
-60

dB

80

dB

-50

Frequency response bandwidth, post output filter, -3 dB

UNIT

85%

Gain

AV

TYP

20

dB
20000

Hz

20

kHz

10

kO

NOTE 2: Output power IS thermally limited, TA = 85°C

electrical characteristics, headphone amplifier, HPVCC
(unless otherwise noted)

=5 V, RL =32 n, TA =25°C, See Figure 3

PARAMETER

TEST CONDITIONS

MIN

Power supply rejection ratio

MAX

UNIT

-10

VN

dB

-60
-1

Uncompensated gain range
IDD

Supply current

9

12

rnA

IDD(MUTE)

Supply current, mute mode

9

12

rnA

IDDIS/D\

Supply current, shutdown mode

20

30

I1A

= 5V, RL = 32 n, gain set at -10VN,

operating characteristics, headphone amplifier, HPVcc
TA 25°C, See Figure 3 (unless otherwise noted)

=

PARAMETER
Po

TEST CONDITIONS

Output power

THD=0.5%,
1= 1 kHz

Crosstalk

1=1 kHz

Frequency response bandwidth, post output lilter, -3 dB
BOM

Maximum output power bandwidth

ZI

Input impedance

MIN

TYP

MAX

-60

dB
20

kHz

20

kHz

>1

PARAMETER

UNIT
mW

50

20

operating characteristics, HPREG 5-V regulator, TA

t

TYP

MO

=25°C (unless otherwise noted)

TEST CONDITIONS

Vo

Output voltage

VDD = PVDD = LPVDD = RPVDD = 8 V to 14 V,
10=Ot090mA

lOS

Short-circuit output current

VDD = PVDD = LPVDD = RPVDD = 8 V to 14 vt

MIN
4.5
90

TYP

MAX
5.5

UNIT
V
rnA

Pulse width must be limited to prevent exceeding the maximum operating virtual junction temperature 01 150°C.

thermal shutdown
TEST CONDITIONS

PARAMETER
Thennal shutdown temperature
Thennal shutdown hysteresis

MIN

TYP

MAX

UNIT

165

°C

30

°C

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2-147

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION
r---------------~
FAUUO~

,

I

1

FAULT1~

I

~

I

HPREG
I SHUTDOWN
HPREG ~ MUTE

I 141S
LOUTN'

~MODE

I

I,

2F
0.2J.1h

I

O.22J.1F

-=l=-

i

12 V 9,16 ,I LPVDD
1 J.1F
Balanced
Differential
Input Signal

I
'1

1S J.1H

~---.----,
40

-::-

LOUTP 1011

,

{-1~,

LINP

I,

---.J ~ LINN
~'1
1 J.1F
-11 LCOMP

I 29
V2PSn

6
r -_ _ _..:c

r431l

I

1000 pF -L
T

1000PFT

-::-

...L

1

!

Balancad
Differential
Input Signal

1

HPLOUT~

COSC

HPDR~

I
1 J.1F
I
44 I

1 J.1F

1

RINN

1

1

33,34 ! RPVDD
7,20,46:47 1AGND
112,13,27,3637 PGND

-::-

.

1
I

1

12 V _ _....._--"'21'-'-'2""'8'-\ PVDD

SOOkn

"Ii
o
HPREG

-

1

0.1 J.1F

ROUTN !-"'34"'35"'--fY"'v'Y"\...-_ _..-_--,

1

1
I
I
1

1L _ _ _ _ _ _ _ _ _ _ _ _ _ROUTP
3839
_._ ~1
I
Figure 1. 12-V, 4-0 Test Circuit

~TEXAS
2-148

l

I

HPLIN

~HPRIN
I
-32
---*-.--""'--!I
HPVCC
100kn

:::L
T47nF

VCP ...",,22,-----.
I
I
T

iI

~

0.1 J.1F

CP2Ir2",-s_-,-

12 V

.I

To HPVCC

I
HPDLI 23
.CP1 1
..."",24,-----.

{ -1~IRINP
---.J~
~\I

rT. . . .-

t-n-

HPROUT
HPREG!.....!.!17---.

!

-::-

T1J.1F
_

VDD~ 1~V

!

~I

1000 pF T

I
I

RCOMP

.

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

40

TPA032D04
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

r---------------,

HPREG
HPREG

FAULTO~

I
I
I
~
I

I

FAULT1~
SHUTDOWN

~ MUTE

I4
-=I

I

I 1415
LOUTN'

II

1 JlF

I

I

LPVDD

I
I
I 29

t-Lli

nI

J

V2P5n

I J

RCOMP

VDD~

1000 PFT

I
~I COSC

-.l

HPROUT
HPREG 1-11.!..!7--~It--- To HPVCC

--1~
1 IlF

12 V

33,34
7,20,46,47
112,13,27,36,37

I

~

HPDL 123
.-CP1 1-1"",24,------,

I

i

I

RPVDD
AGND
PGND

VCP 1--1",,22,------,

I
I

II

12V _ ____.----"2"'1!.!:2"'--SI PVDD

~

-=-

I
-l.-T 47 nF
CP2I:-""25'--__....J

RINN

I

I

T 0,11lF

HPDR~

I
I
44 I
--1~1 RINP
11lF

{

1;V

t--M--

I!

-=Balanced
Differential
Input SIgnal

F
11l

I
HPLOUT~

-=-

1000 pF T

ao

LOUTP ......,..,10,,-,1,,-1-f'rYY~_ _ _ _ _- - - '

Balanced {--1
L1NP
Differential
JLU LINN
Input Slgnal
~~
11lF
r -_ _ _...::6'-11 LCOMP

1000PF~

' - * - -.......----,

I
I

MODE

12 V9,16

30llH

HPLIN

l

T

0.11lF

-=-

ROUTN !--"=34",35"'-fY'<'Y\.._._--_._---,

I
I
~HPRIN
I
I
I
_ _ _~~-~3=-2 I HPV
I
I
CC
I
I
ROUTP 3839
100kn
IL _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ ~I
500kO

To
HPREG

ao

-=Figure 2, 12-V, 8-0 Test Circuit

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2-149

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S203A - DECEMBER 1999 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION
r---------------~
---.!.;
SHUTDOWN
142
2
FAULTOt-=-

HPREG

I

HPREG ~ MUTE
HPREG ~! MODE

~

FAULT1 1

1
LPVDD
1

1 14,15
LOUTN ~

vl!.!!...f

12

5

-: - n

L

, LlNP

LOUTP

LiNN

1
. -_ _----"6'-; LCOMP

~

~ RCOMP

-::- 1000pF T

.1

--

48

470 PFT

-::-

L
-

-

44

±

1

!

33,34

COSC

HPLOUT!18

1

HPROUTI 31

I RINP

HPREG

1

1

123

CP1 1 24

1

12V~ PVDD

Right SE
HP Input

---1

1
125

J..- 47 nF
T
_

1

l

CP2 i-',=-----'
VCP ....,12=2_--.

1
l

---1

1

i

T 0.111F

1
1
1
HPLIN
1
1
ROUTN 134,35
1
1 HPRIN
11
1
1 HPVCC
1
1
ROUTP 138,39
1L _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ J1

PTrl
19

100 kn

100 kn

0.111F 100

kn

VDD
500 kn
To
HPREG

HPROUT
---.>+--------

30
32

0.1

!1F

Figure 3. Headphone Test Circuit

~lExAs
2-150

11---*----.,

HPVCC

HPDLi-=---------'

1

Left SE
HP Input

1.-.!4

3211F

HPDR ....,2=6_ _---'

1
11 RPVDD

100 knl

111F

-

VDDr- 12V

!

7:20:46:47 ! AGND
12,13_27_36_37 _ PGND

HPLOUT

J-.L

18

~ RINN

-::12V

!

V2P5~

1
11

'

J-!!!.ll-

11

-.L 1

1000 pF T

1

INSTRUMENTS

POST OFFICEOOX 655303 • DALLAS, TEXAS 75265

320

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
HPREG
To System
Control
100kn

{~;H-;;~;;;;----------'1OO"~
2

.....

~

MUTE
+---'\Mt~_ _ _--",3,--!1 MODE
916 I
12 V ---4+t--::t:
.......--::t:-+-''''''
'-"-II LPVDD
10 IlF -L -:r:- 1 IlF -:r:- 1 IlF I

~

V

11lF
LeftClass-DBalanced
Differential Input
Signal

FAULTO t--==:....--+-.......- } To System
I 41
FAULT1 t-'-''------.--Control
I

I

V

I

I

LOUTN

{---l~

LlNP

---l~

LINN

11lF

I

~

I
-=I
.--...:48",,-!1 COSC

Right Class-D Balanced
Differantial Input
Signal

---l~
11lF

12 V

II

MODE

220l1F

HPROUT I 31
RINP

I

HPREG}-IT- HPVCC

RINN

HPDR;-I2:>6:....-_ _ _---.

I

::t:33,34 I RPVDD
10 F-L -:r:- 11lF -:r:- 11lF I
V

100kO

I

+::t:

Il~

HPVcc

HPLOUT~ f---"'2=20'-!:I1::...F--4......._ _~

I

{---l ~

40
0.221lF

-::;r

II

-=- 1 IlF

0.22"Fh
..

I
151lH
V2P51-'12""9'----_ _--,
18
12V4111.F
VDDI
::t:
I 1 I1F

~I

-=-1OO0 PF r

1000 pF rL

I
I

LOUTP ...,..,10,,-,1,,-1-"ryy~-----.-----,

43 I RCOMP

T

14,15

I

.--_ _ _--"6'-f1 LCOMP

1000 pF

1ookO

142

1 kn

I

V

1 kn

7 20 46 47 AGND
121327 36 37 I PGND

I

:!:.

12 V

21,28 I PVDD

1 IlF ~ HPLOUT
O.lIlF

Left SE
HP Input
RlghtSE
HP Input

---1

100 kO

I
I

19

1--""-_-,t

l

!-==-"ryy~-----.--J O.lI1F

HPLIN

100 kn

I

lookO

I

47nF

---1 f--'V\II.,.......>----,30=---!1 HPRiN

VDD

40

O.lI1F 100
kn

32 I HPVCC
ROUTP J-38""""3",-9-,,ryy~_ _---._----,

I
HPROUT
To -41....- -..
HPREG

T

IL. _ _ _ _ _ _ _ _ _ _ _ _ _ _ _ .JI

1511.H

0.11lF
NOTE A.

~ =power ground and ~

=analog ground

Figure 4. TPA032D04 Typical Configuration Application Circuit

-!!1

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

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TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SL0S203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and ZI, the TPA032004's input resistance forms a
high-pass filter with the corner frequency determined in equation 8.

fC(highpass)

=

2:rt~ICI

(8)

ZI is nominally 10 k.Q

The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where the specification calls for a flat bass response down to 40 Hz. Equation 8 is
reconfigured as equation 9.

C I = _1_

(9)

2:rtZ I fc

In this example, CI is 0040 IiF so one would likely choose a value in the range of OA7liF to 1 IiF. A low-leakage
tantalum or ceramic capacitor is the best choice for the input capacitors. When polarized capacitors are used,
the positive side of the capacitor should face the amplifier input, as the dc level there is held at 1.5 V, which is
likely higher than the source dc level. Please note that it is important to confirm the capacitor polarity in the
application.
differential input
The TPA032004 has differential inputs to minimize distortion at the input to the IC. Since these inputs nominally
sit at 1.5 V, dc-blocking capacitors are required on each of the four input terminals. If the signal source is
Single-ended, optimal performance is achieved by treating the signal ground as a signal. In other words,
reference the signal ground at the signal source, and run a trace to the dc-blocking capacitor, which should be
located physically close to the TPA032004. If this is not feasible, it is still necessary to locally ground the unused
input terminal through a dc-blocking capacitor.
power supply decoupling, Cs
The TPA032004 is a high-performance Class-O CMOS audio amplifier that requires adequate power supply
decoupling to ensure the output total harmoniC distortion (THO) is as low as possible. Power supply decoupling
also prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling
is achieved by using two capacitors of different types that target different types of noise on the power supply
leads. For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-seriesresistance (ESR) ceramic capacitor, typically 0.1 IiF placed as close as possible to the device's various Voo
leads, works best. For filtering lower-frequency noise Signals, a larger aluminum electrolytic capacitor of 10 IiF
or greater placed near the audio power amplifier is recommended.
The TPA032004 has several different power supply terminals. This was done to isolate the noise resulting from
high-current switching from the sensitive analog circuitry inside the IC.

~TEXAS

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INSTRUMENTS
POST OFFICE BOX 655303 • OALLAS. TEXAS 75265

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
mute and shutdown modes
The TPA032D04 employs both a mute and a shutdown mode of operation designed to reduce supply current,
100, to the absolute minimum level during periods of non-use for battery-power conservation. The SHUTDOWN
input terminal should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN
low causes the outputs to mute and the amplifier to enter a low-current state, 100 = 20 JlA. Mute mode alone
reduces 100 to 10 mA.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

output filter components
The output inductors are key elements in the performance of the class-D audio amplifier system. It is.important
that these inductors have a high enough current rating and a relatively constant inductance over frequency and
temperature. The current rating should be higher than the expected maximum current to avoid magnetically
saturating the inductor. When saturation occurs, the inductor loses its functionality and looks like a short circuit
to the PWM signal, which increases the harmonic distortion considerably.
A shielded inductor may be required if the class-D amplifier is placed in an EMI sensitive system; however, the
switching frequency is low for EMI considerations and should not be an issue in most systems. The dc series
resistance of the inductor should be low to minimize losses due to power dissipation in the inductor, which
reduces the efficiency of the circuit.
Capacitors are important in attenuating the switching frequency and high frequency noise, and in supplying
some of the current to the load. It is best to use capacitors with low equivalent-series-resistance (ESR). A low
ESR means that less power is dissipated in the capacitor as it shunts the high-frequency signals. Placing these
capacitors in parallel also parallels their ESR, effectively reducing the overall ESR value. The voltage rating is
also important, and, as a rule of thumb, should be 2 to 3 times the maximum rms voltage expected to allow for
high peak voltages and transient spikes. These output filter capacitors should be stable over temperature since
large currents flow through them.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

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TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

efficiency of class-D vs linear operation
Amplifier efficiency is defined as the ratio of output power delivered to the load to power drawn from the supply.
In the efficiency equation below, PL is power across the load and Psup is the supply power.
Efficiency

P

= 11 = __L_
Psup

A high-efficiency amplifier has a number of advantages over one with lower efficiency. One of these advantages
is a lower power requirement for a given output, which translates into less waste heat that must be removed
from the device, smaller power supply required, and increased battery life.
Audio power amplifier systems have traditionally used linear amplifiers, which are well known for being
inefficient. Class-O amplifiers were developed as a means to increase the efficiency of audio power amplifier
systems.
'
A linear amplifier is designed to act as a variable resistor network between the power supply and the load. The
transistors operate in their linear region and voltage that is dropped across the transistors (in their role as
variable resistors) is lost as heat, particularly in the output transistors.
The output transistors of a class-O amplifier switch from full OFF to full ON (saturated) and then back again,
spending very little time in the linear region in between. As a result, very little power is lost to heat because the
transistors are not operated in their linear region. If the transistors have a low on-resistance, little voltage is
dropped across them, further reducing losses. The ideal class-O amplifier is 100% effiCient, which assumes that
both the on-resistance (rOS(on) and the switching times of the output transistors are zero.
the ideal class-D amplifier

To illustrate how the output transistors of a class-O amplifier operate, a half-bridge application is examined first
(see Figure 5).
VDD

+

Figure 5. Half-Bridge Class-D Output Stage

Figures 6 and 7 show the currents and voltages of the half-bridge circuit. When transistor M1 is on and M2 is
off, the inductor current is approximately equal to the supply current. When M2 switches on and M1 switches
off, the supply current drops to zero, but the inductor keeps the inductor current from dropping. The additional
inductor current is flowing through M2 from ground. This means that VA (the voltage at the drain of M2, as shown
in Figure 5) transitions between the supply voltage and slightly below ground. The inductor and capacitor form
a low-pass filter, which makes the output current equal to the average of the inductor current. The low pass filter
averages VA, which makes VOUT equal to the supply voltage multiplied by the duty cycle.

~TEXAS

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INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA032D04
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SL0S203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
the ideal class-O amplifier (continued)
Control logic is used to adjust the output power, and both transistors are never on at the same time. If the output
voltage is rising, M1 is on for a longer period of time than M2.
Inductor Current

o~---+--~--~----~--~--~--~--~-----*

M1 onl M1 offl M1 ani
M2 offl M2 on I M2 offl • • •
Time

Figure 6. Class-O Currents

~--~--~---r--~----r---'---~--~----VDD

VOUT

M1 on IM1 off IM1 ani
M20fflM2on IM20ffl···
Time

Figure 7. Class-O Voltages

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

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TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
the ideal class-O amplifier (continued)
Given these plots, the efficiency of the class-O device can be calculated and compared to an ideal linear
amplifier device. In the derivation below, a sine wave of peak voltage (Vp) is the output from an ideal class-O
and linear amplifier and the efficiency is calculated.
LINEAR

CLASS-O
Vp
VL(rms) =
A

Vp

!2

VL(rms)

I ) = IL(rms)Vx VL(rms)
verage (00
00

P

_
L-

V

= !2
L(rms)
RL

2

V 2

= _P_

2 RL

2 Vp
Average (100 ) = it x R
L
Psup

P

= Voo

x Average(loo)

Psup

Voox IL(rms) x VL(rms)
Voo

- -----'-;-,---'-----'----'-

SUP -

= Voo

Efficiency

=

x Average ( 100 )

=

Voo Vp
2
R
x it
L

PL
11 = - Psup
V p2

Efficiency

Efficiency

PL

\ P sup

2RL
Efficiency = 11 = Voo x - - 2 Vp
-x1t
RL

=1

Efficiency

= 11 = - -

=

11

=

11

1t

=-

4

Vp
x -VOO

=

In the ideal efficiency equations, assume that Vp Voo, which is the maximum sine wave magnitude without
clipping. Then, the highest efficiency that a linear amplifier can have without clipping is 78.5%. A class-O
amplifier, however, can ideally have an efficiency of 100% at all power levels.
The derivation above applies to an H-bridge as well as a half-bridge. An H-bridge requires approximately twice
the supply current but only requires half the supply voltage to achieve the same output power-factors that
cancel in the efficiency calculation. The H-bridge circuit is shown in Figure 8.

Voo

Voo
+ vOUTL

L
Figure 8. H-Bridge Class-O Output Stage

~TEXAS

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INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA032D04
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
losses in a real-world class-D amplifier
Losses make class-O amplifiers nonideal, and reduce the efficiency below 100%. These losses are due to the
output transistors having a nonzero r08(on), and rise and fall times that are greater than zero.
The loss due to a nonzero r08(on) is called conduction loss, and is the power lost in the output transistors at
nonswitching times, when the transistor is on (saturated). Any r08(on) above 0 n causes conduction loss.
Figure 9 shows an H-bridge output circuit simplified for conduction loss analysis and can be used to determine
new efficiencies with conduction losses included.
VDD= 12V

rDS(on)

0.36Q

5 MQ

rDS(off)

0.36Q

rDS(on)

4Q
rDS(off)

5 MQ

Figure 9. Output Transistor Simplification for Conduction Loss calculation
The power supplied, P8UPo is determined to be the power outputto the load plus the power lost in the transistors,
assuming that there are always two transistors on.
Efficiency =

1]

=

1]

Efficiency

PL

=-P8UP

12 2r 08(on)

Efficiency =

RL

1]

2r 08(on)
Efficiency =
Efficiency

=

1]

1]

+ 12RL

+

RL

n, RL = 4 n)
= 85% (at all output levels r 08(on) = 0.36 n, RL = 4 n)
= 95%

(at all output levels r 08(on)

= 0.1

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

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TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
losses in a real-world class-D amplifier (continued)
Losses due to rise and fall times are called switching losses. A diagram of the output, showing switching losses,
is shown in Figure 10.

H

tswon

+

tSWoff

tsw

Figure 10. Output Switching Losses
Rise and fall times are greater than zero for several reasons. One is that the output transistors cannot switch
instantaneously because (assuming a MOSFET) the channel from drain to source requires a specific period
of time to form. Another is that transistor gate-source capacitance and parasitic resistance in traces form RC
time constants that also increase rise and fall times.
Switching losses are constant at all output power levels, which means that switching losses can be ignored at
high power levels in most cases. At low power levels, however, switching losses must be taken into account
when calculating efficiency. Switching losses are dominated by conduction losses at the high output powers,
but should be considered at low powers. The switching losses are automatically taken into account if you
consider the quiescent current with the output filter and load.
class-D effect on power supply
Efficiency calculations are an important factor for proper power supply design in amplifier systems. Table 2
shows Class-D efficiency at a range of output power levels (per channel) with a 1-kHz sine wave input. The
maximum power supply draw from a stereo 10-W per channel audio system with 4-0 loads and a 12-V supply
is almost 26 W. A similar linear amplifier such as the TPA032D04 has a maximum draw of greater than 50 W
under the same circumstances.
Table 2. Efficiency vs Output Power in 12-V 4-0 H-Brldge Systems
Output Power (W)

Efficiency (%)

Peak Voltage (V)

Internal Dissipation (W)

0.5

41.7

2

0.7

2

66.7

4

1.0

5

75.1

6.32

1.66

8

78

8

2.26

10

77.9

8.94t

2.84

t High peak voltages cause the THD to increase

~TEXAS

2-158

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
class-O effect on power supply (continued)
There is a minor power supply savings with a class-O amplifier versus a linear amplifier when amplifying sine
waves. The difference is much larger when the amplifier is used strictly for music. This is because music has
much lower RMS output power levels, given the same peak output power (see Figure 11); and although linear
devices are relatively efficient at high RMS output levels, they are very inefficient at mid-to-Iow RMS power
levels. The standard method of comparing the peak power to RMS power for a given signal is crest factor, whose
equation is shown below. The lower RMS power for a set peak power results in a higher crest factor
Crest Factor

=

10 log

PPK

Pnns

Time

Figure 11. Audio Signal Showing Peak and RMS Power

~TEXAS

INSTRUMENTS
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2-159

TPA032D04
10-W STEREO CLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA032D04 data sheet, one can see that when the
TPA032D04 is operating from a 12-V supply into a 4-0 speaker that 20-W peaks are available. Converting watts
to dB:
P

dB

=

10Log

(:w)

=

10Log

ref

(~O) = 6 dB

(17)

Subtracting the crest factor restriction to obtain the average listening level without distortion yields:
6.0 dB - 18 dB
6.0 dB - 15 dB

- 12 dB (15 dB crest factor)

= -

9 dB (15 dB crest factor)

6.0 dB - 12 dB = - 6 dB (12 dB crest factor)
6.0 dB - 9 dB = - 3 dB (9 dB crest factor)
6.0 dB - 6 dB = - 0 dB (6 dB crest factor)
6.0 dB - 3 dB = 3 dB (3 dB crest factor)
Converting dB back into watts:

Pw

=

1O PdB / 10 x P

ref

(18)

= 315 mW (18 dB crest factor)

= 630 mW (15 dB crest factor)

=

1.25 W (12 dB crest factor)

=

2.5 W (9 dB crest factor)

= 5 W (6 dB crest factor)

=

10 W (3 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 10 W of continuous power output with a 3 dB
crest factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 12-V, 4-0 system, the internal dissipation in the
TPA032D04 and maximum ambient temperatures are shown in Table 3.

~TEXAS

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INSTRUMENTS
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TPA032D04
10·W STEREO CLASS·D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

crest factor and thermal considerations (continued)
Table 3. TPA032D04 Power Rating, 12-V, 4-0, Stereo
MAXIMUM AMBIENT
TEMPERATURE

PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

20

10W(3dB)

2.84

23°C

20

5W(6dB)

1.66

75°C

20

2.5W(9dB)

1.12

100°C

20

1.25 W (12 dB)

0.87

111°C

20

630 mW (15 dB)

0.7

118°C

20

315mW(18dB)

0.6

123°C

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
data from the dissipation rating table, the derating factor for the DCA package with 6.9 in 2 of copper area on
a multilayer PCB is 44.8 mW/oC. Converting this to 9JA:

e

JA

1
Derating

=

(19)

=_1_
0.0448

=

22.3°C/W

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given 9JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA032D04
is 150°C. The internal dissipation figures are taken from the Efficiency vs Output Power graphs.
T A Max

= T J Max - e JA P D

(20)

150 - 22.3(0.7 x 2) = 118°C (15 dB crest factor)
150 - 22.3(2.84 x 2) = 23°C (3dB crest factor)
NOTE:
Internal dissipation of 1.4 W is estimated for a 1O-W system with a 15 dB crest factor per channel.

The TPA032D04 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. Table 3 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-0
speakers dramatically increases the thermal performance by increasing amplifier efficiency.

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2-161

TPA032D04
10-W STEREOCLASS-D AUDIO POWER AMPLIFIER
SLOS203A - DECEMBER 1999 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced DCA package is based on the 56-pin TSSOP, but includes a thermal pad (see Figure 12)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine~pitch, surface-mount package can
be reliably achieved.

Side View (8)

End View (b)

Bottom View (c)

Figure 12. Views of Thermally Enhanced DCA Package

~TEXAS

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INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-1

Contents
Page
TPA152

75-mW Stereo Audio Power Amplifier ......................... 3-3

TPA102

150-mW Stereo Audio Power Amplifier ....................... 3-17

TPA112

150-mW Stereo Audio Power Amplifier ....................... 3-39

TPA122

150-mW Stereo Audio Power Amplifier ........................ 3-63

0

TPA302

300-mW Stereo Audio Power Amplifier ....................... 3-85 .

Q)
t/)
t/)

TPA301

350-mW Stereo Audio Power Amplifier ...................... 3-105

TPA311

350-mW Stereo Audio Power Amplifier ...................... 3-125

l>

TPA701

700-mW Stereo Audio Power Amplifier ...................... 3-155

m

TPA711

700-mW Stereo Audio Power Amplifier ...................... 3-175

l>

TPA721

700-mW Stereo Audio Power Amplifier ...................... 3-205

C.
0
"tJ
0

TPA4860

1-W Stereo Audio Power Amplifier .......................... 3-225

I

_.

C

TPA4861

1-W Stereo Audio Power Amplifier .......................... 3-249

TPA0253

1-W Mono Audio Power Amplifier ........................... 3-271

TPA0103

1.75-W Three-Channel Audio Power Amplifier ............... 3-277

...

TPA0102

2-W Stereo Audio Power Amplifier .......................... 3-313

TPA0112

2-W Stereo Audio Power Amplifier .......................... 3-349

3

TPA0122

2-W Stereo Audio Power Amplifier .......................... 3-381

TPA0132

2-W Stereo Audio Power Amplifier .......................... 3-413

:e
(I)

l>

-....__..

"C

TPA0142

2-W Stereo Audio Power Amplifier .......................... 3-441

(I)

TPA0152

2-W Stereo Audio Power Amplifier .......................... 3-469

t/)

TPA0162

2-W Stereo Audio Power Amplifier .......................... 3-497

TPA0202

2-W Stereo Audio Power Amplifier .......................... 3-525

TPA0212

2-W Stereo Audio Power Amplifier .......................... 3-565

TPA0213

2-W Mono Audio Power Amplifier ........................... 3-597

TPA0222 '

2-W Stereo Audio Power Amplifier .......................... 3-607

TPA0223

2-W Mono Audio Power Amplifier ........................... 3-639

TPA0232

2-W Stereo Audio Power Amplifier .......................... 3-643

TPA0233

2-W Mono Audio Power Amplifier ........................... 3-671

TPA0242

2-W Stereo Audio Power Amplifier .......................... 3-675

TPA0243

2-W Mono Audio Power Amplifier ........................... 3-703
6-W Stereo Audio Power Amplifier .......................... 3-707

...

TPA1517

3-2

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
•
•
•
•
•
•
•
•

o PACKAGE

High-Fidelity Line-Out/HP Driver
75-mW Stereo Output
PC Power Supply Compatible
Pop Reduction Circuitry
Internal Mid-Rail Generation
Thermal and Short-Circuit Protection
Surface-Mount Packaging
Pin Compatible With TPA302

(TOP VIEW)

IN1GND

V01
MUTE

BYPASS
IN2-

VDD

Vo2

description
The TPA 152 is a stereo audio power amplifier capable of less than 0.1 % THD+N at 1 kHz when delivering
75 mW per channel into a 32-0 load. THD+N is less than 0.2% across the audio band of 20 to 20 kHz. For
10 kQ loads, the THD+N performance is better than 0.005% at 1 kHz, and less than 0.01 % across the audio
band of 20 to 20 kHz.
The TPA 152 is ideal for use as an output buffer for the audio CODEC in PC systems. It is also excellent for use
where a high-performance head phonelline-out amplifier is needed. Depop circuitry is integrated to reduce
transients during power up, power down, and mute mode.
Amplifier gain is externally configured by means of two resistors per input channel and does not require external
compensation for settings of 1 to 10. The TPA 152 is packaged in the B-pin SOIC (D) package that reduces board
space and facilitates automated assembly.

typical application circuit
RF

voo
Stereo Audio
Input

-=-=-

Rr
-=CI

RL

From System
Control

RL

L~ ~I
RF

~~~:o~: sl=~r:.i;;~:::,::,c:=:=-

standard warranty. PrO~UOIlon processing dOlI not necessarily include
testing of all parameters.

~TEXAS

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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

-=-

Copyright © 2000, Texas Instruments Incorporated

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A - JUNE 199B - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICE
TA
-40°C to 85°C

SMALL OUTLINE
TPA1520t

t The 0 packages are available taped and reeled. To
order a taped and reeled part, add the suffix R
(e.g., TPAI520R)

Terminal Functions
TERMINAL
NAME

NO.

BYPASS

3

1/0

DESCRIPTION
BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected to a O.I-I1F
to l-I1F capacitor.

GND

7

IN1-

8

I

IN2-

4

I

IN2- is the inverting input for channel 2.

MUTE

2

I

A logic high puts the device into MUTE mode.

GNO is the ground connection.
IN1- is the inverting input for channell.

VOO

6

I

VOO is the supply voltage terminal.

VOl

1

0

VOl is the audio output for channell.

V02

5

0

VQ2 is the audio output for channell.

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TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A - JUNE 1998 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)*
Supply voltage, VDD ....................................................................... 6 V
Input voltage, VI ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. -{l.3 V to VDD + 0.3 V
Continuous total power dissipation ..................... internally limited (See Dissipation Rating Table)
Operating junction temperature range, TJ .......................................... -40°C to 150° C
Operating case temperature range, T C ............................................ -40°C to 125° C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
t Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

D

DERATING FACTOR
724mW

5.8mWFC

464mW

376mW

recommended operating conditions
Supply voltage, VDD
Operating free-air temperature, TA

dc electrical characteristics at TA

MIN

MAX

4.5

5.5

V

-40

85

°C

TYP

MAX

=25°C, Voo =5 V

PARAMETER

TEST CONDITIONS

MIN

Output offset voltage

VOO

UNIT

UNIT

10

mV

rnA

Supply ripple rejection ratio

VDD = 4.9 V to 5.1 V

81

IDD

Supply current

See Figure 13

5.5

14

IDD(MUTE)

Supply current in MUTE

5.5

14

ZI

Input impedance

>1

dB

rnA
MO

ac operating characteristics Voo = 5 V, TA = 25°C, RL = 32 n (unless otherwise noted)
PARAMETER
Po
THD+N
BOM

Vn

TEST CONDITIONS

Output power (each channel)

THD S 0.03%,

Gain = 1,

Total harmonic distortion plus noise

Po=75mW,
See Figure 2

20 Hz-20 kHz, Gain = 1.

Maximum output power bandwidth

AV=5,

THD20

kHz

80°
See Figure 12

Supply ripple rejection ratio

1 kHz,

65

dB

Mute attenuation

See Figure 15

110

dB

ChICh output separation

See Figure 13

102

dB

Signal-to-Noise ratio

Vo = 1 V(rms),

Noise output voltage

See Figure 10

Gain = 1

See Figure 11

104
6

dB
ILV(rms)

t Measured at 1 kHz.
NOTES: 1. The dc output voltage is approximately VDot2.
2. Output power is measured at the output pins of the IC at 1 kHz.

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3-5

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A - JUNE 1998 - REVISED MARCH 2000

ac operating characteristics Voo

=5 V, TA =25°C, RL =10 kQ

PARAMETER

THD+N

BOM

kSVR

Vn

t

TEST CONDITIONS

MIN

TYP

VI = 1 V(nns),
See Figure 6

20 Hz-20 kHz, Gain = 1,

VO(pp)=4V,
See Figure 8

20 Hz-20 kHz, Gain = 1,

Maximum output power bandwidth

G=5,

THD <0.02%, See Figure 6

>20

Phase margin

Open loop,

See Figure 16

80°

Supply voltage rejection ratio

1 kHz,

CB=1I1F,

Mute attenuation

See Figure 15

Total harmonic distortion plus noise

ChICh output separation

See Figure 13

Signal-to-Noise ratio

Vo = 1 V(nns),

Noise output voltage

See Figure 10

Gain = 1,

See Figure 12

See Figure 11

MAX

UNIT

0.005%
0.005%
kHz

65

dB

110

dB

102

dB

104
6

dB
I1V(rms)

Measured at 1 kHz.

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
THD+N

Total harmonic distortion plus noise

vs Output power

THD+N

Total harmonic distortion plus noise

vs Frequency

THD+N

Total harmonic distortion plus noise

vs Output voltage

Vn

Output noise voltage

vs Frequency

SNR

Signal-to-noise ratio

vs Gain

11

Supply ripple rejection ratio

vs Frequency

12
13,14

1,4
2,3,6,8,9
5, 7
10

Crosstalk

vs Frequency

Mute Attenuation

vs Frequency

15

Open-loop gain and phase

vs Frequency

16, 17

Closed-loop gain and phase

vs Frequency

18

IDD

Supply current

vs Supply voltage

19

Po

Output power

vs Load resistance

20

PD

Power dissipation

vs Output power

21

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TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A - JUNE 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

.,.

vs

OUTPUT POWER

FREQUENCY

2

..

E

I

CD

I

f= 1 kHz
AV=-1 VN

1v~ ~~~I~V

0
0

~u

0.01

-

0

.E

V

"-

0.1

C

/

E

::t

:e{:.

I

0.01

v/
~

...

f'.,
Av=-1VN =

I

Z
+

Z

+

Q

i!=

lVI=I~~~

I I 11

c

'E

0.1

'c0

:e{:.

..
+

0

.

PO=75mW
RL=320

z

c

::t

2

I

'0

Z
+

~u

.,.
CD

'0

'f0

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

Q

::t

I-

0.001

0.001
1

10

20

30

40

50

60

70

80

90

20

100

Po - Output Power - mW

1k

10k 20k

f - Frequency - Hz

Figure 1

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER
2

~ RL=320

20kHz
0.1
1""0

.........

1't-:

1 kHz

0.01
20Hz -

100

1k

10k 20k

0.001
0.1

f - Frequency - Hz

10

100

Po - Output Power - mW

Figure 3

Figure 4

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3-7

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A JUNE 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT VOLTAGE

FREQUENCY

2

0.1

'#.

'#.

f = 1 kHz
Ay=-1 Y/v
RL = 10 kO

..
I

GI

'0

YO=1 Y(rms)
RL = 10 k.Q

I
GI

.!!!
0

z

Z

+

+
c

c
0

1:
0

~

0

1:

0.1

~

..

..

c0

C

Ay=-2Y/v

Ii

~

0.01

S

11I1111

0.01

0

II

Ii

:c

Ay=-5Y/v

0

:c

S

~I

Ay=-1 Y/v

(:.
I

Z

'"""

+

Q

:c

Z

+

r-

Q

:c

I-

0.001

I-

o

0.001
0.2

0.4

0.6

0.8

1.2

1.4

1.6

1.8

20

100

Yo - Output Yoltage - Y(rms)

Figure 5

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT VOLTAGE

FREQUENCY

2

0.1

~

I

.~

51

"0

z

+

+

f=20kHz -

0

c

--r-

0.1

0

1:
0

]i

..

Q

c0

~

E
01

S

0.01

..........

'02

0.01

Ii
s
~

f=2~~Z

~

:c

~
I
Z

I'

I"

I

+

Z

+

~

Q

~

YO(pp)=4Y
Ay=-1 Y/v
RL = 10 k.Q

I

c

:c

'"

'#.

Ay =-1 Y/v
RL= 10kO

0
Z

:e0
~

10k 20k

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

'#.

1k
f - Frequency - Hz

Q

:c

f=1kHz
0.001
0.1

I-

0.001
0.2

0.4

2

20

YO - Output Yoltage - Y(rms)

Figure 7

100

1k
f - Frequency - Hz

FigureS

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10k 20k

TPA152
75·mW STEREO AUDIO POWER AMPLIFIER
SLOS210A - JUNE 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

OUTPUT NOISE VOLTAGE

vs

vs

FREQUENCY

FREQUENCY

0.1

20

'#.

VI = 1 V(rms)
AV=-l VN

I

3l
·0
z

~

+

c

.s

III
Dl

:!

RL=32D.j V

III

is
£
c

I

II

~

~

-I~

CD

0.01

.!!

0

E
III

10

VOO=5V
BW = 10 Hz to 22 kHz
RL = 32 D. to 10 kD.
AV=-l VN

I......

::t

"iii

~I

RL = 10,47, and 100 kn

-

-

z0

-

0

:i

t

I

:f'

Z
+
C
::t

I-

0.001
20

100

lk

1
20

10k 20k

100

f - Frequency - Hz

lk

Figure 9

Figure 10

SIGNAL-TO-NOISE RATIO

SUPPLY RIPPLE REJECTION RATIO

vs

vs

GAIN

FREQUENCY

110

0
I

VOO=5V
RL = 32 D. to 10 kn

RI =20kD.
-10
105
III
"tJ

I
0

:;

100

a:

.~
~

ic

r

III

~
\

95

"tJ

90

~ !:::::-..J

RL=10kD.

RL=3~F=====

a:

z

-30

0

-40

l

CD

-50

~

a.
a.

-60 I-

a:

-70

a.
a.

-60

:g

~

85

r--.. ........

~ CB=O.lIlF

ia:
c

i'..

lill

-20

I

I

UJ

10k 20k

f - Frequency - Hz

~

=

r--.1'

I'

f\..

.....

CB=lIlF

l"-

UJ

-

1

2

3

4

5

6

7

8

9

10

r---

~

.L

V

CB=2.5V

-90
80

'"

r-....

-100
20

Gain-VN

J
100

lk

10k 20k

f - Frequency - Hz

Figure 11

Figure 12

-!!1 TEXAS

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3-9

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A - JUNE 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-80

-70

III

"

-80

-80

PO=75mW
VOO=5V
RL=32n
CB=lIlF
AV=-l VN

"r-o.

-70

-

-80
III

"I

I

1'"

-90

1e'"

f'I...

"1\

S

-100

"

Right to Left

~

0

:;:

Jl

~

-110

100

lk

-100

-130
20

10k 20k

I'~

Right t~ Left

1\

1'\

rv
100

lk

Figure 14
vs

FREQUENCY

GI

"c

V~OI=51V
t-

RL=32n
CB=l IlF

90

I

0

-100

~

..

-110

::e

-120

i:::I
C

.l!!
:::I

,

-130

-140
20

100

lk

f - Frequency - Hz

Figure 15

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10k 20k

;7'
,L

Left to Right

Figure 13·

-80

_

~

f - Frequency - Hz

MUTE ATTENUATION

3-10

II

f - Frequency - Hz

-70

-

~

-120

I 1111111

-120

-90

-110

Left to Right

2Q

VO=l V
VOO=5V
RL= 10kn
CB = 1 IlF
AV=-l VN

10k 20k

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A-JUNE 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN·LOOP GAIN AND PHASE

vs
FREQUENCY
100
No Load

I'

140

SO

m
'C

J

I'

120

60

c

'iii
CI

a.
0

40

...

"

I-

.9
CD

a.

100

60

"

0

-20
100

1k

10k

..

II>

.c

Q.

"

20

0

J
III

SO

c

0

160

100k

1M

40

\

~
10M

20
0
100M

f - Frequency - Hz

Figure 16
CLOSED·LOOP GAIN AND PHASE

vs
FREQUENCY
1S5
O.S
1S0

0.6

m

'C

/'f'

0.4

175

J

c

0.2

a.

0

'iii
CI
0
0

\

0

J

170

...J

III

II>

.!

Q.

,; -{l.2
CD
II>

165

0

0 -{l.4

RI=20kO
Rf=20kO
RL=320
CI= 1 IlF
AV=-1 VN

-{l.6
-{l.S
-1

10

100

1k

10k

100k

160

155
1M

f - Frequency - Hz

Figure 17

~TEXAS

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3-11

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A - JUNE 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE
vs
FREQUENCY
185
0.8
180

0.6
III

IL

0.4

'C

1\

I

c
1i'i

0.2

a.

0

~..

-0.2

175
0

CJ

~

..
!
I

0
0

170

III

1:1.

165

-0.4

RI=20kQ
Rf=20kQ
RL=10kQ
CI= 11lF
Av=-1 VN

-0.6
-0.8

160

1111

-1

10

100

1k
10k
f - Frequency - Hz

155
1M

100k

Figure 18
SUPPLY CURRENT
vs
SUPPLY VOLTAGE

CC
E

OUTPUT POWER
vs
LOAD RESISTANCE

10

90

9

80
~

8

I

C

~
:::I

I

7

~

0

~

a.
a.

-

6

:::I

III
I

c
c

5

'5a.
'5

0

I

~

4

60

50
4Q

\

\
~

~

30
20

3
4.5

5

5.5

10
30

50

VDD - Supply Voltage - V

70

~ ...........
90

--r-- t--

Figure 20

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

-

110 130 150 170 190 210

RL - Load Resistance - Q

Figure 19

3-12

-

70

E

I

THD+N=0.1%
AV=-1 V/V

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A-JUNE 1998- REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION

vs
OUTPUT POWER

100

I
RL=32f.!

80
c

!

60

~

I
I
C
D.

-----

r--

/

I

V

~

40

20

o

o

5

15

10

20

25

Po - Output Power - mW

Figure 21

APPLICATION INFORMATION

selection of components
Figure 22 is a schematic diagram of a typical application circuit.

Audio Input 1

-

CI
111F

RF
20kf.!

RI
20kQ

Shutdown
(from System Control)

2

V01

IN1-

MUTE

GND

Rot
20kf.!

8

-=-

7

Rct
100f.!

-=-

111F
3
CB
111F
CI
111F
Audio Input 2

---1

T
-=-

4

IN2

VDD

IN2-

V02

6

VDD

5

RI
20kf.!
RF
20kQ

RCt
100f.!

-=-

-=-

t These resistors are optional. Adding these resistors improves the depop performance of the TPA 152.

Figure 22. TPA152 Typical Application Circuit

~TEXAS

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3-13

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS21 OA - JUNE 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
gain setting resistors, RF and RI
The gain for the TPA 152 is set by resistors RF and RI according to equation 1.
Gain = -

(~~)

(1 )

Given that the TPA 152 is a MOS amplifier, the input impedance is very high, consequently input leakage
currents are not generally a concern although noise in the circuit increases as the value of RF increases. In
addition, a certain range of RF values are required for proper start-up operation of the amplifier. Taken together
it is recommended that the effective impedance seen by the inverting node of the amplifier be set between 5
kil and 20 kil. The effective impedance is calculated in equation 2.

RR
F+

Effective Impedance = R F ~

(2)
I

As an example, consider an input resistance of 20 kil and a feedback resistor of 20 kil. The gain of the amplifier
would be -1 and the effective impedance at the inverting terminal would be 10 kil, which is within the
recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kil; the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF' This, in effect, oreates a
low-pass filter network with the cutoff frequency defined in equation 3.
f

1
c(lowpass) - 2:n;R F CF

(3)

For example if RF is 100 kil and CF is 5 pF then fco(lowpass) is 318 kHz, which is well outside the audio range.

input capacitor, C.
In the typical application, an input capacitor, CI> is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and R, form a high-pass filter with the corner frequency
determined in equation 4.
f

1
c(highpass) - 2:n;R I C ,

(4)

The value of C, is important to consider as it directly affects the bass (lOW frequency) performance of the circuit.
Consider the example where R, is 20 kQ and the specification calls for a flat bass response down to 20 Hz.
Equation 4 is reconfigured as equation 5.
C,

=

1

(5)

2:n;R , fC(highpass)

In this example, C, is 0.40 IlF, so one would likely choose a value in the range of 0.47 IlF to 1 IlF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI> C,) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage atthe inputto the amplifier
that reduces useful headroom, especially in high-gain applications (> 10). For this reason a low-leakage
tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the
capacitor should face the amplifier input in most applications, as the dc level there is held at VDoI2, which is
likely higher that the source dc level. Please note that it is important to confirm the capacitor polarity in the
application.
'

~TEXAS

3-14

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SLOS210A-JUNE 1998- REVISED MARCH 2000

APPLICATION INFORMATION
power supply decoupling, Cs
The TPA 152 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure that the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 ~F, placed as close as possible to the device VDD lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 ~F or greater placed near
the power amplifier is recommended.

mid rail bypass capacitor, CB
The midrail bypass capacitor, CB, serves several important functions. During startup or recovery from shutdown
mode, CB determines the rate at which the amplifier starts up. This helps to push the start-up pop noise into
the subaudible range (so slow it can not be heard). The second function is to reduce noise produced by the
power supply caused by coupling into the output drive signal. This noise is from the midrail generation circuit
internal to the amplifier. The capacitor is fed from a 160-k.Q source inside the amplifier. To keep the start-up pop
as low as pOSSible, the relationship shown in equation 6 should be maintained.

1
(C B x 160

s_1_

kU)

(6)

(CIR I)

As an example, conSider a circuit where CB is 1 ~F, CI is 1 ~F and RI is 20 kO. Inserting these values into the
equation 9 results in:
6.25

s

50

which satisfies the rule. Bypass capacitor, CB, values of 0.1 ~F to 1 ~F ceramic or tantalum low-ESR capaCitors
are recommended for the best THO and noise performance.

output coupling capacitor, Cc
In the typical single-supply single-ended (SE) configuration, an output coupling capacitor (Cd is required to
block the dc bias at the output of the amplifier thus preventing dc currents in the load. As with the input coupling
capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by
equation 7.

(7)
The main disadvantage, from a performance standpoint, is that the load impedances are typically small, which
drive the low-frequency corner higher. Large values of Cc are required to pass low frequencies into the load.
Consider the example where a Cc of 68 ~F is chosen and loads vary from 320 to 47 kO. Table 1 summarizes
the frequency response characteristics of each configuration.

~TEXAS

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3-15

TPA152
75-mW STEREO AUDIO POWER AMPLIFIER
SlOS210A- JUNE 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode
RL

Cc

LOWEST FREQUENCY

320

6811F

73Hz

10,0000

6811F

0.23 Hz

47,0000

6811F

0.05 Hz

As Table 1 indicates, headphone response is adequate and drive into line level inputs (a home stereo for
example) is very good.
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following
relationship:
(8)

output pull-down resistor, RC + RO
Placing a 100-0 resistor, Re, from the output side of the coupling capacitor to ground insures the coupling
capacitor, Ce, is charged before a plug is inserted into the jack. Without this resistor, the coupling capacitor
would charge rapidly upon insertion of a plug, leading to an audible pop in the headphones.
Placing a 20-kO resistor, Ro, from the output of the Ie to ground insures that the coupling capacitor fully
discharges at power down. If the supply is rapidly cycled without this capacitor, a small pop may be audible in
10-kO loads.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real capacitor can be modeled
simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the
beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the
real capacitor behaves like an ideal capacitor.

~TEXAS

INSTRUMENTS
3-16

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA102
150·mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

UGN PACKAGE

• 150 mW Stereo Output
• PC Power Supply Compatible
- Fully Specified for 3.3 V and 5 V
Operation
- Operation to 2.5 V
•
•
•
•

(TOP VIEW)

BYPASS

IN1-

GND

Vo1

SHUTDOWN

VDD

V02

Pop Reduction Circuitry
Internal Mid-Rail Generation
Thermal and Short-Circuit Protection
Surface-Mount Packaging
- PowerPADTM MSOP

• Pin Compatible With LM4881

description
The TPA 102 is a stereo audio power amplifier packaged in an 8-pin PowerPADTM MSOP package capable of
delivering 150 mW of continuous RMS power per channel into 8-0 loads. Amplifier gain is externally configured
by means of two resistors per input channel and does not require external compensation for settings of 1 to 10.
THD+N when driving an 8-0 load from 5 V is 0.1 % at 1 kHz, and less than 2% across the audio band of 20 Hz
to 20 kHz. For 32-0 loads, the THD+N is reduced to less than 0.06% at 1 kHz, and is less than 1% across the
audio band of 20 Hz to 20 kHz. For 1O-kQ loads, the THD+N performance is 0.01 % at 1 kHz, and less than 0.02%
across the audio band of 20 Hz to 20 kHz.

typical application circuit

325kn
RF

325kn

VDD 6
Voo

l-

ie

S

VDD/2

-=

Audio
Input
R,

~e,
IL

8

IN1-

1

BYPASS

4

IN 2-

'I

7

r+

V02 5

I

ec

~C

T

RI

~

-=-

...A

I

FromShutdown
Control Clrcuit

3

<

A

Vo1

CB.l.

Audio
Input

~

r+

I
I

SHUTDOWN

I

'~C
Bias
Control

I~
2

-

RF

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

~~: 9'='::t~~s';'~:~:::Ie:'::~":;

standard wananty. Production processing does not necessarily include
testing of .11 paramelerS.

~TEXAS

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POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3-17

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICE
MSOpt

MSOP
Symbolization

TPA102DGN

TIAAC

TA
-40°C to 85°C

tThe DGN package IS available In left-ended tape and reel only (e.g.,
TPA 102DGNR).

Terminal Functions
TERMINAL
NAME
BYPASS

NO.

110

DESCRIPTION

I

Tap to voltage divider for internal mid-supply bias supply. Connect to a 0.1 !iF to 1 !iF low ESR capacitor for
best performance.

1

GND

2

I

GND is the ground connection.

IN1-

8

I

IN1- is the inverting input for channell.

IN2-

4

I

IN2- is the inverting input for channel 2.

SHUTDOWN

3

I

Puts the device in a low quiescent current mode when held high.

VDD

6

I

VDD is the supply voltage terminal.

V01

7

0

Vo 1 is the audio output for channell.

V02

5

0

V02 is the audio output for channel 2.

absolute maximum ratings over operating free-air temperature (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ........................................................... -0.3 V to Voo + 0.3 V
Continuous total power dissipation ................................................ internally limited
Operating junction temperature range, T J .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for -10 seconds ............................... 260°C
t Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
DGN

TA;!;25°C
POWER RATING
2.14

wt

DERATING FACTOR
ABOVE TA 25°C

TA = 70°C
POWER RATING

TA = 85°C
POWER RATING

17.1 mW/oC

1.37W

1.11W

=

:I: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMAOO2), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions.
Supply voltage, VDD
Operating free-air temperature, TA

~TEXAS

INSTRUMENTS
3-18

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

MIN

MAX

2.5

5.5

UNIT
V

-40

85

°C

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

de electrical characteristics at TA = 25°C, VDD = 3.3 V
PARAMETER

VIO

Input offset voltage

PSRR

Power supply rejection ratio

IDD

Supply current

IDD(SD)
ZI

TEST CONDITIONS

MIN

TYP

UNIT

5

mV

1.5

3

rnA

Supply current in SHUTDOWN mode

10

50

Input impedance

>1

ac operating characteristics, VDD

83

VDD = 3.2 V to 3.4 V

dB
~

MQ

=3.3 V, TA =25°C, RL =8 Q

PARAMETER

TEST CONDITIONS

MIN

TYP

Po

Output power (each channel)

THD:50.1%

THD+N

Total harmonic distortion + noise

Po =70 mW,

20-20 kHz

2%

Maximum output power BW

G= 10,

THD<5%

>20

Phase margin

Open loop

BOM

MAX

MAX

70t

UNIT

mW
kHz

58°

Supply ripple rejection ratio

f = 1 kHz

68

Channel/channel output separation

f = 1 kHz

86

dB

SNR

Signal-to-noise ratio

PO=100mW

100

dB

Vn

Noise output voltage

9.5

IiV(rms)

dB

t Measured at 1 kHz

de electrical characteristics at TA = 25°C, VDD = 5 V
TEST CONDITIONS·

PARAMETER

VIO

Input offset voltage

PSRR

Power supply rejection ratio

IDD

Supply current

MIN

TYP

mV

1.5

3

rnA

100

76

VDD = 4.9 Vto 5.1 V

UNIT

5

IDD(SD)

Supply current in SHUTDOWN mode

60

ZI

Input impedance

>1

ac operating characteristics, VDD

MAX

dB
~

MQ

=5 V, TA =25°C, RL =8 Q

PARAMETER

TEST CONDITIONS

MIN

TYP

Po

Output power (each channel)

THD:50.1%

70t

THD+N

Total harmonic distortion + nOise

PO=150mW,

20-20 kHz

2%

BOM

Maximum output power BW

G = 10,

THD<5%

>20

Phase margin

Open loop

MAX

UNIT

mW
kHz

56°

Supply ripple rejection ratio

f= 1 kHz

68

dB

Channel/Channel output separation

f= 1 kHz

86

dB

SNR

Signal-to-noise ratio

PO=150mW

Vn

Noise output voltage

100

dB

9.5

IiV(rms)

t Measured at 1 kHz

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-19

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

ac operating characteristics, Voo

=3.3 V, TA =25°C, RL =32 Q

PARAMETER

TEST CONDITIONS

Po

Output power (each channel)

THDSO.1%

THD+N

Total harmonic distortion + noise

PO=30mW,

2~20kHz

BOM

Maximum output power BW

AV=10,

THO <2%

Phase margin

Open loop

MIN

TYP

MAX

40t

UNIT

mW

0.5%
kHz

>20

58°

Supply ripple rejection ratio

f = 1 kHz

68

dB

ChanneVchannel output separation

f= 1 kHz

97

dB

SNR

Signal-to-noise ratio

PO=1OOmW

Vn

Noise output voltage

100

dB

9.5

I1V(rms)

t Measured at 1 kHz

ac operating characteristics, Voo

=5 V, TA =25°C, RL =32 Q

PARAMETER

TEST CONDITIONS

MIN

TYP

40t

Po

Output power (each channel)

THDSO.1%

THD+N

Total harmonic distortion + noise

PO=60mW,

2~20kHz

BOM

Maximum output power BW

AV = 10,

THO <2%

Phase margin

Open loop

MAX

UNIT

mW

0.4%
>20

kHz

56°
dB

Supply ripple rejection ratio

f= 1 kHz

68

ChanneVchannel output separation

f= 1 'kHz

97

dB

SNR

Signal-to-noise ratio

PO=150mW

100

dB

Vn

Noise output voltage

9.5

I1V(rms)

t Measured at 1 kHz

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE

THD+N

Total harmonic distortion plus noise

vs Frequency
vs Power output

Vn

1,2,4,5,7,8,
10,11,13,14,
16,17,34,36
3,6,9,
12,15,18

Power supply rejection ratio

vs Frequency

19,20

Output noise voltage

vs Frequency

21,22

Crosstalk

vs Frequency

23-26,37,38

Mute attenuation

vs Frequency

27,28

vs Frequency

29,30

Open-loop gain
Phase margin
Output power

vs Load resistance

31,32

100

Supply current

vs Supply voltage

33

SNR

Signal-to-noise ratio

vs Voltage gain

35

Closed-loop gain
Phase
Power dissipation

vs Frequency

39-44

vs Output power

45,46

~TEXAS

INSTRUMENTS
3-20

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

#.

10

VOO=3.3V
PO=30mW
Ce=1I1F
RL=32n

I

3:
"0
z

+
c

.
I

~

·0

L

lv 1;'111J I~IV

.!o!
c

AV=-1VIV
RL=32n
Ce = 1l1 F

CD

Z

+

AV=-5~1V

Ic

E VOO=3.3V

#.

c
0

V'

'E

~

V'

V V

0.1

.~

0

PO=15mW

0.1

0

i

::c
iii

;§

~

0.01

..!!!

I"
AV=-1 VIV

S
~

I

~

PO=10mW

::c

1"'8

I-

0.01

I

z

Z

c+

~
~

::c

....

0.001
20

100

1k

10k 20k

PO=30mW

III

0.001
20

Jill
lk

100

f - Frequency - Hz

Figure 1

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

10

10

#.

VOO=3.3V
RL=32n
AV=-1 VIV
Ce= 1l1F

I

.~

Z

+
c

0

#.

VOO=5V
Po=60mW
RL=32n
Ce=1I1F

I

.~

Z

+

27 kHZ

10kHz

'E

c

~

.s

AV=-10VN
I I

.s

II>

is

.!t!

.2
c

.2

.

0

c

c

0

!!!
::c

10k 20k

f - Frequency - Hz

0.1

S

~

--

I
Z

+
C
::c

....

0.01

JOHz

.L

-.

S

~

J

10

,.

~

0.01

--

./

L

l.Y V
IL

I

Z

~
~

I..--'

1

0.1

!!!
!

~kHz

AV =-5 VN

50

Av=-1 VN

I I III

0.001
20

Po - Output Power - mW

100

1k

10k 20k

f - Frequency - Hz

Figure 3

Figure 4

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-21

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C -AUGUST 1998

REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER
10

10

#.

VOO=5V
RL=32Q
AV=-1 VN
CB= 1 f.1F

I

·1z

~

Z

+
c

~0

-=

0.1

0

Ii

PO=30mW

!§.. Po=15mW

!

~

c0

::t:
Oi

0.1

rr---.

I

Z

+
::t:

+
::t:

Po=60mW

Q

0.001
20

Q

II

III

I-

100

I-

1k

10k 20k

20Hz

~

0.01
0.002

0.01

Figure 5

vs

FREQUENCY

FREQUENCY
10

VOO=3.3V
RL=10kQ
Po= 1oof.1F
CB= 1 f.1F

z

#.

1
~

+

+

c

~0

..

C

~0

'Iii
is

AV=-5VN
0.1

.~

0

Ii

~

0.1

0

::t:

!

VOO=3.3V
RL=10kQ
AV=-1 VN
CB=1 flF

I

c

~

I

V

0.01

t-

t-

Oi

;2

I

0.01

PO=45f.1W

I~

!"""i"

./

I

z

Z

+
::t:

AV=-2VN

Q

+
Q
::t:

-

I-

r-

I-

0.001
20

100

II ""10k

1k

0.001
20k

20

f - Frequency - Hz

Po= 9O f.1W

Po = 130f.1W

IIII

I I III

100

1k
f - Frequency - Hz

Figure 7

FigureS

~TEXAS

INSTRUMENTS
3-22

0.2

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

a
is

0.1

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

I

II

Po - Output Power - W

f - Frequency - Hz

#.

r- ....

1 kHz

;2

"'"

-

-

.

is

Ii

~

1 I:::

0.01

I
Z

10kHz

~

~

::t:

20, kHz

~

~

'Iii
is

..

Voo=5V
r=AV=-1VN
r- RL=32Q
_CB=1f.1F

I

+
c

C

r=

#.

POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

II ""10k

20k

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

VOO=3.3V
RL=10kn
AV=-1 VN
CB = 111F

tft
I
GI
III

'0

Z
+
C
0

'E
0

~

"
';:

0.1

E
:I!
]i
~

20Hz _

-

0

0.01

~

I

Z
+
Q

=

1-

~1'I.-!i~~~~~AiV~=~-5~V~N~II~

~

0.1

I
1 kHz
I

0.001

AV=-2VN

I I

10

5

AV= 1 VN

~~ill~llllLlt.ii"""'~II~~

0.01

20Hz

:I:
~

10 kHz

1\

100

200

L-I....J....L..U.J.U....----l--L-Wu.JJUJ...---L---'-...J.,.J..........1J---J

20

100

Po - Output Power - I1W

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10

10

tft

VOO=5V
RL=10kn
AV=-1 VN
CB= 111F

I
GI

.!!

z0

+

tft

j
+

c

c
o

'E

.e

~

III

Q

I

Po = 200 I1W ~

"-

"0.01

is

Po = 3OOl1W

0.1

0

~

VOO=5V
RL=10kQ
AV=-1 VN
CB=1I1 F

I

~

]i

10k 20k

Figure 10

TOTAL HARMONIC DISTORTION PLUS NOISE

C

1k
f - Frequency - Hz

Figure 9

u

r---

j I IIII

0.001

I"~

f':iII

I~

I

Z

+

Q

I 1111111

0.001
20

§
:I:
'"

20 Hz
20k~z

]i

].....I
10kHz 1 kli'z- -

Q

j:

II

100

I II

0.001
1k

\

0.01

7
~

Po = 100 l1W

j:

0.1

u

C

10k 20k

5

10

f - Frequency - Hz

100

I 1-

500

Po - Output Power - I1W

Figure 11

Figure 12

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-23

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 199B - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

;I.

..
z
..
a:

2

I

CD

~

'0

0

;:

j

~

0.1

10
;I.

J0

V

AV=

'"
/

......

u

z

Po=30mW

+
c

~~

~

AV=-

L

.......

Po=15mW

~

0.1

.2

c

~i"'"

0

:I:

!

~

0.01

~
I

I

Z
+
C
:I:

Z

~

Po=75mW

III
II :1111

I-

0.001
20

100

1k

0.001

10k 20k

20

f - Frequency - Hz

vs
FREQUENCY

I

2~kHZ

10kHz

..

a:'"

:l

,~

~

A~~

0

a

i

1 kHz
0.1

!

~I

0.1

"'"

Av=-

t--

IV

~

~~

0.01

{!.

20Hz

I

~

aLI

:I:

I-

I-

0.01
10m

0.1

0.3

0.001
20

100

1k
f - Frequency - Hz

Po - Output Power - W

Figure 16

Figure 15

3-24

v~:..ti'V1V

c

:e0

!

Z
+
C
:I:

=JI ~~~=Jv~.

~

'c0

~

Z

--

.s
'"

is

F

.~

r•

_1- 11IIIIII
VOO=5V .
PO=100inW
~ RL=8Q
t- CB= 111F

E

CD

I-

~

2

;I.

~AV=-1 VIV

+
c

TOTAL HARMONIC DISTORTION PLUS NOISE

OUTPUT POWER
Voo = 3.3 V
t- RL=8Q

.!z

10k 20k

vs
10
I

1k

Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

;I.

100

f - Frequency - Hz

Figure 13

:I:

,

~

~

L~

0.01

-

is

IV

{!.

i!:

Voo = 3.3 V
RL=8Q
AV=-1 VIV

I

~

'co

i!

-

IIIIII I I
:Ay = -2 VN.

t-

c

FREQUENCY

J II

1=

:l

vs

FREQUENCY

~o~~~~~~

PO=75mW
RL=8Q
t-CB=1I1F

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

-!11
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

POWER OUTPUT

10

'#.

10

1= VOO=5V

..

'#.

.!!!
0
Z
+

I

'0

+
c

0

~

.!:!
c
0

E
til

E
til

....."""

.....
0.01

:c

t--

I

Z

+

+

PO=10mW

Q

:c

100

Q

20Hz

0.01
10m

10k 20k

0.1

f - Frequency - Hz

Po - Output Power - W

Figure 17

Figure 18

SUPPLY RIPPLE REJECTION RATIO

SUPPLY RIPPLE REJECTION RATIO

vs

vs

FREQUENCY

FREQUENCY

0

0

-10
III
'0
I

c

0

~
'iii'
II:

.!!
Q.

f

~

Q.
Q.

::J
II)

~

-'I

'""'"

i!:

I111111
1k

0.001

?Hz

I I

0.1

;§

Z

I-

10kHz

r---

'ii

I

20

-

~

~~

E Po=60mW

0.1

:c

'!

~

~0

'E0

~u

I khkHz

z

Po=30mW

c

'c0

VOO=5V
RL=8f.l
AV=-1 VN

...

i=_AV=-1VN
RL =8f.l

I

VOO=3.3V
RL = 8 f.l to 10 kf.l

-I'

I I

-20

~

-30

~~

-40

1'"

-50
-60

C

B=

I' ~

-90

-100

-

20

III
'0
I

~

'"

~

-40 ~!"o

is.
Q.

-50

k:iI

v

~W

Ii:

-30

~
Q.

-60 -'-;'B=- f.1

::J
II)

-70

r--....

1k

I'

I~ I

I"

I'..

r": ~N

10k 20k

-100

20

f - Frequency - Hz

I
.!,

= 1 f.1F

t'....

1']\
~

k&

V

R\~

-90

100

B= .1 tlF

-aD

VI

r-

VOO=5V
RL = 8 f.l to 10 kf.l -

r-- ...

.
l

..

r--....

"~
I"'"

-20

c

I

Ib(1lF

. JB1=111I11
-70
ypasrl= 1.65
-a0

-10

Bypass = 2.5 V

1-1111111
100

I

1k

10k 20k

1- Frequency - Hz

Figure 19

Figure 20

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-25

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

OUTPUT NOISE VOLTAGE

OUTPUT NOISE VOLTAGE

vs

vs

FREQUENCY

FREQUENCY

20

20

.,
S

,
~

.,
~

10

=[

I

10

I

;

III

~

~

Iz

.~

Z

:;

:;

~

~

0

0

I

I

VOO=3.3V
BW = 10 Hz to 22 kHz
AV=-1 VN
RL = 8 0 to 10 kO

>C
1
20

100

VOO=5V
BW = 10 Hz to 22 kHz
RL = 8 0 to 10 kO
AV=-1 VN

::f
1k

1 ~'LLUlll
100
20

10k 20k

f - Frequency - Hz

I

I I I I

1k

Figure 21

Figure 22

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-60

-50
PO=25mW
VOO=3.3V
RL=320
CB= 111F
AV=-1 VN

-65
-70
-75
ID
'1:1
I

~

e

0

-60

N2 OOU

V

-65

-95

"-

-100

~

~

1/

V

ID
'1:1
I

~

e'"
0

-75

INi T

-90

bi U~ii-

1k

10k 20k

IN2TOOUT

'"

-60

-65

I III
100

-70

L """/

-105
-110
20

,Po' ~ ;'00 m~

-55 r- VOO=3.3V
RL=80
-60 r- CB=1I1F
-65 r- AV=-1 VN

!'\

-90

~~

1V
to-

fo"'"

IN1 TO OUT 2

-95
-100

20

f - Frequency - Hz

100

1k
f - Frequency - Hz

Figure 24

Figure 23

~TEXAS

INSTRUMENTS
3-26

10k 20k

f - Frequency - Hz

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

10k 20k

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C-AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-60

-50
VOO=5V
PO=25mW
CB= 1 i-LF
RL=32n
AV=-1 VN

-65
-65
-75
III

"...
I

;..

e

(.)

-80

I '~

VOO=5V
-55 t- PO= 100 mW
CB = 1 i-LF
-60 -RL=Sn
AV=-1 VN
-65

-

III

V

I'

-85

r-

IN2TO?UT1

-90

~

-95

r'"

-100

>
~

~

"...
I..
I

e

(.)

-70

-....~r-.

-80
-85

~~

IN2TOf\UT1

-75

l'

J..;I-"

INI1TillM -95

III

-110
20

11111

100
1k
f - Frequency - Hz

-100
20

10k 20k

Figure 25

vs

FREQUENCY

III

"cI

FREQUENCY

,

-30

III

-40

"c

ii:::I

-50

ii:::I

~

-60

0

C
III

.!!
:::I

::&

-30

I

0

-40

!

-50

::&

-70

.!!
:::I

-70

-60

-80

-80

-90

-90

-100
20

10k 20k

MUTE ATTENUATION

vs

VOO = 3.3 V
-10 I- RL=32n
CB=1 i-LF
-20

1111111
1k
100
f - Frequency - Hz

Figure 26

MUTE ATTENUATION
0

-

-90

IN1 TO OUT 2
-105

7

100

1k
f - Frequency - Hz

10k 20k

-100
20

Figure 27

1k
100
f - Frequency - Hz

10k 20k

Figure 28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-27

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE MARGIN

vs
FREQUENCY
100

No Load

~~

80

~

III
'C
I

V~~~13:3V

r-...

0
0

...t!:

..

a.
0

i'
40

Gain ~~

20

120°

ilthlL

60

c
'iii
CJ
a.

150°
I

c

90°

"E'

:I

'"

:I

60°

f.
I

....E

30°

,,~
0

-20

0°

10

100

lk

-300
10M

lOOk

10k

f"'" Frequency - Hz

Figure 29
OPEN-LOOP GAIN AND PHASE MARGIN

vs
FREQUENCY
100

~~~~II~VI

i'
80
III
'C
I

c

1\

60

iii

No Load

"

CJ

a.
0

-i!.
0

II~haU
Gain

t

lk

10k

lOOk

..

60°

=
f.

300

'£

I

~

0

1M

f - Frequency - Hz

Figure 30

~TEXAS

3-28

90°

,~

20

100

120°

~

,~

40

-20

150°
I

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

-30°
10M

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

vs

vs

LOAD RESISTANCE

LOAD RESISTANCE

120

100

300

1\

\

;:

E
I

80

~
0

D..

'5
I

rP

~

"'-

40

i\

!
!

200

D.. 150

~

THO~N=l ~

250 ~

;:

"

60

f

0

THO+N=l %
VOO = 3.3 V
AV=-l VIV
-

VOO=5V
Av=-l VIV

r-..

'5

.............

I

I"-.....

""-

100

rP

)00..

20

50

o

0
16

8

24

32

40

48

56

64

8

16

RL - Load Resistance - f.l

24

c(

--

vs

SUPPLY VOLTAGE

FREQUENCY

#.

t- RL = 10 kf.l
t- CB =lIlF

!I

ii:

6

'E

0.1

i

0.8

.~

0.6

j

:::I

-

64

~ AV=-l VN

~

(.)

III
I
Q
Q

56

r= VI = 1 V

I

11

I

Q.
Q.

48

TOTAL HARMONIC DISTORTION PLUS NOISE

C

~

40

vs

E

§

32

r--

Figure 32

SUPPLY CURRENT

1.2

I"-.....I--

RL - Load Resistance - f.l

Figure 31

1.4

-

~

0.4

lii

0.2

Z

~

0.01

~

....

I

C!i

0

J:
I- 0.001

2.5

3

3.5

4

4.5

5

5.5

20

100

1k

10k 20k

f - Frequency - Hz

VOO - Supply Voltage - V

Figure 33

Figure 34

-!!I TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-29

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998

REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SIGNAL-TQ..NOISE RATIO
104

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

VOLTAGE GAIN

FREQUENCY

"#.

I

102 ~

ID

"I
0

~

100

0

98

,;j
z0

.:ras

\

I

Dl

=

-CB=1I1F

is:

~

'\

I

~

0.1 _ _ _

,g

"

:--...

96

I

II:

Z

'"

~ Av=-1 VN
r- RL 10 kO -1--I--t+-H+tl----t-t-+t-t+Ht----1

rg

c

iii

~ Voo=5V

I

VI=1 V

94

I'--.~

j

!

~ 0.01

.........

I

f'...

92

~

::t:

0.001

I-

1

2

3

4
5
6
7
8
AV - Voltage Gain - VN

9

10

~~~II~~;I~!mll~
L-....l-.I....l..u..LW---'--'-w..J..wJ.._"-I...........u.J.Lo...--J

20

100

CROSSTALK

~

r-

-80

r-

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-60

~o~ ~ 131.~1 ~

-100

..

~

e

VO=1 V
RL=10kO
CB= 111F

~

. . . . . r-.

(J

-120

1'-1/

-130

1I

V

"'r0-

--

"I

t?"

IIII

--90

ID

i..
e

~~

IN2toOUT1

-110

!

11111111
VoO=5V
-70 t-- Vo = 1 V
RL = 10 kO
-80 t-- CB=1I1F

-90

ID

"I

-70

-100

.........

-110

r---

-150
20

100

-120

'l

I 'I-

-130

10k 20k

III! I III

-150
20

100

1k
f - Frequency - Hz

Figure 38

Figure 37

~TEXAS

INSTRUMENTS
3-30

I--'~
I""
IN1 toOUT2

-140

1k
f - Frequency - Hz

~~

IN2toOUT1

(J

"'"

11 1

~

IIII

IN1 toOUT2
-140

10k 20k

Figure 36

Figure 35

-60

1k
f - Frequency - Hz

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED·LOOP GAIN AND PHASE

vs
FREQUENCY

2000

11111111

1800

Phase

1/

1600

1\

CII

1400

=
a.

.c

1200

J

VOO=3.3V
RI=20kn
RF=20kQ
RL=32Q
CI= 111F
AV=-1 VN

...
III
I

30

~

20

~

10

c

800

.Ill!llil

Q.

1
o

1000

1111

0

~

1111111

-10
10

100

1k

10k
100k
f - Frequency - Hz

1M

Figure 39
CLOSED·LOOP GAIN AND PHASE

vs
FREQUENCY

2000

11111111

1800

Phase

1/

1600

r"I

1400

!Ii

.c

a.

1200

J
...

VOO=5V
RI=2Okn
RF=20kQ
RL=32Q
CI= 111F
AV=-1 VN

III
I

30

·ii

20

c

"

i
o

800

11111111

10

o

1000

Gain
11111

r

1111111

-10
10

100

1k

10k

100k

1M

f - Frequency - Hz

Figure 40

-!!1 TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-31

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

VB
FREQUENCY

200°

J' HIli

180°

Pha~~"

~

160°

1\

CD

1400

:I

.c

II.

120°
Voo=3.3V
RI=20kO
RF=20kO
RL=80
CI=1I1F
AV=-1 VN

1/

III
'0

I

c

OJ
CJ

40

!

20

J

100°
80°
60°

'ci~~~'
11111111

o

I---'

-20

10

100

10k

1k'

"'"

100k

1M

f - Frequency
- Hz
"

Figure 41
CLOSED·LOOP GAIN AND PHASE

VB
FREQUENCY

"..

/

""'"'

Phase

200°
180°

......

160°
140°
120°

VOO=3.3V
RI=20kO
RF=20kO
RL=10kO
CI=1I1F
AV=-1 VN

:
20
10

o
-10

-

10

80°

11111111
Gain

100

,""'"

1k
10k
100k
f - Frequency - Hz

Figure 42

~TEXAS

3-32

100°

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

1M

..
CD

.!
II.

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S213C - AUGUST 199B - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED·LOOP GAIN AND PHASE

vs
FREQUENCY

200°

11111111

180'

Phase

/

"

11111111

c

~

120'

~

II

.c
II-

100'
SO'
60°

11111111

1c!~~~11

20

i~
Co

140'

VOO=5V
RI=20kO
RF=20kn
RL=SO
CI=1IlF
Av=-1 VN

'I

III
'D
I

160'

~

11111111

~

40°

11111111

V
10

11111111
100

1k
10k
100k
f - Frequency - Hz

1M

Figure 43
CLOSED·LOOP GAIN AND PHASE

vs
FREQUENCY

200'

11111111
-~

Phase

V

30

iii
c:I

20

!

10

U

-10

c

1

160°

..

III
01

11111111

140° .c
II-

111111111

III
'D
I

180°

120°

VOO=5V
RI=20kn
RF=20kn
RL = 10 kn
CI=1IlF
AV=-1 VN

100°
SO°

11111111

11111111
Gain

o

'"'
11111111
10

100

1k

10k

100k

1M

f - Frequency - Hz

Figure 44

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-33

TPA102
1SD-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION/AMPLIFIER
vs
OUTPUT POWER

80

180
VOO=3.3V

70

~

I

8?

~
a.
E

/

40
30

C

20

~

o

20 40

~

E

I

.....

"~
60

E
C

1,,\

80
60
40

"

20

ao 100 120 140 160 1aO

L

100

8?

~
a.

"-

120

I

"- ~

~ ~o

.J 40~

10

140

~

/
1/
V' ~o

abI-"'""

VOO=5V

160

f' I'....

I

50

o

.--

alo

L

60

E
I

POWER DISSIPATION/AMPLIFIER
vs
OUTPUT POWER

200

o

V

./

~

160

-'-

-.... r--.

LV

'L

V

b.... ......

--~r--.....
0

"""

~

02040

Load Power - mW

60

I...........

i'-

80100120140160180
Load Power - mW

Figure 45

Figure 46

~TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

200

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
gain setting resistors, RF and RI
The gain for the TPA 102 is set by resistors RF and RI according to equation 1.
Gain = -

(~~)

(1 )

Given that the TPA 102 is a MOS amplifier, the input impedance is very high. Consequently input leakage
currents are not generally a concern, although noise in the circuit increases as the value of RF increases. In
addition, a certain range of RF values is required for proper start-up operation of the amplifier. Taken together
it is recommended that the effective impedance seen by the inverting node of the amplifier be set between
5 kil and 20 kil. The effective impedance is calculated in equation 2.
R R
Effective Impedance = R F ~
F

+

(2)
I

As an example, consider an input resistance of 20 kil and a feedback resistor of 20 kil. The gain of the amplifier
would be -1 and the effective impedance at the inverting terminal would be 10 kil, which is within the
recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kil, the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF- This, in effect, creates a
low-pass filter network with the cutoff frequency defined in equation 3.
f

1
c(lowpass) - 2nR F C F

(3)

For example, if RF is 100 kil and CF is 5 pF then fc(lowpass) is 318 kHz, which is well outside the audio range.
input capacitor, CI
In the typical application, an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 4.
f

1
c(highpass) - 2nR I C I

(4)

The value of CI is important to consider, as it directly affects the bass (lOW frequency) performance of the circuit.
Consider the example where RI is 20 kil and the specification calls for a flat bass response down to 20 Hz.
Equation 4 is reconfigured as equation 5.
CJ =

1
2nRI fC(highpass)

(5)

In this example, CI is 0.40 IlF, so one would likely choose a value in the range of 0.47 IlF to 1 IlF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI' CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage atthe input to the amplifier
that reduces useful headroom, especially in high-gain applications (>10). For this reason a low-leakage
tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the
capacitor should face the amplifier input in most applications, as the dc level there is held at Vool2, which is
likely higher than the source dc level. It is important to confirm the capacitor polarity in the application.

~TEXAS

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3-35

TPA102
1-50-mW STEREO AUDIO POWER AMPLIFIER
SLOS213C - AUGUSl' 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
power supply decoupling, Cs
The TPA 102 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure that the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
-prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients; spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 I1F, placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals,a larger aluminum electrolytiC capacitor of 10 I1F or greater placed near
the power amplifier is recommended.

midrail bypass capacitor, CB
The midrail bypass capacitor, Cs, serves several important functions. During startup, Cs determines the rate
at which the amplifier starts up. This helps to push the start-up pop noise into the subaudible range (so low it
can not be heard). The second function is to reduce noise produced by the power supply caused by coupling
int01he output drive signal. This noise is from the midrail generation circuit internal to the amplifier. The capacitor
is fed from a 160-kO source inside the amplifier. To keep the start-up pop as low as possible, the relationship
shown in equation 6 should be maintained.
1

(C s

<_1_

x 160

kn) -

(6)

(CIR I)

As an example, consider a circuit where Cs is 1 I1F, CI is 1 I1F, and RI is 20 kn. Inserting these values into the
equation 9 results in: 6.25:S; 50 which satisfies the rule. Bypass capacitor, Cs, values of 0.1 I1F to 1 I1F ceramic
or tantalum low-ESR capacitors are recommended for the best THO and noise performance.

output coupling capacitor, Cc
In the typical single-supply single-ended (SE) configuration, an output coupling capacitor (Cc) is required to
block the dc bias at the output oUhe amplifier, thus preventing dc currents in the load. As with the input coupling
capacitor, .the output coupling capacitor and impedance of the load form a high-pass filter governed by
equation 7.
fc

=

1

(7)

21tRL Cc

The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the
low-frequency corner higher. Large values of Cc are required to pass low frequencies into the load. Consider
the example where a Cc of 68 I1F is chosen and loads vary from 32 n to 47 kn. Table 1 summarizes the
frequency response characteristics of each configuration.

~TEXAS

3-36

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

TPA102
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S213C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 1. Common Load Impedances Vs Low Frequency Output Characteristics In SE Mode

Cc

Lowest Frequency

RL
320

68\!F

73Hz

10,0000

68\!F

0.23 Hz

47,0000

68\!F

0.05 Hz

As Table 1 indicates, headphone response is adequate and drive into line level inputs (a home stereo for
example) is very good.
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following
relationship:

(8)

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this application. A real capacitor can be modeled simply as
a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects
of the capacitor in the circuit. The lower the equivalent value of this reSistance, the more the real capacitor
behaves like an ideal capacitor.

6-Y versus 3.3-Y operation
The TPA 102 was designed for operation over a supply range of 2.5 V to 5.5 V. This data sheet provides full
specifications for 5-V and 3.3-V operation since these are considered to be the two most common standard
voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain
setting, or stability. Supply current is slightly reduced from 3.5 mA (typical) to 2.5 rnA (typical). The most
important consideration is that of output power. Each amplifier in the TPA102 can produce a maximum voltage
swing of VOO -1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as opposed
when VO(PP) = 4 V while operating at 5 V. The reduced voltage swing subsequently reduces maximum output
power into the load before distortion begins to become significant.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75255

~7

3-38

TPA112
150·mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

D OR DGN PACKAGE
(TOP VIEW)

• 150-mW Stereo Output
• Wide Range of Supply Voltages
- Fully Specified for 3.3 V and 5 V
Operation
- Operational From 2.5 V to 5.5 V
• Thermal and Short-Circuit Protection
• Surface Mount Packaging
- PowerPADTM MSOP

V01

VDD

IN1IN1+
GND

V02
IN2IN2+

- sOle
• Standard Operational Amplifier Pinout

description
The TPA 112 is a stereo audio power amplifier packaged in an 8-pin PowerPADTM MSOP package capable of
delivering 150 mW of continuous RMS power per channel into 8-0 loads. Amplifier gain is externally configured
by means of two resistors per input channel and does not require external compensation for settings of 1 to 10.
THD+N when driving an 8-0 load from 5 V is 0.1 % at 1 kHz, and less than 2% across the audio band of 20 Hz
to 20 kHz. For 32-0 loads, the THD+N is reduced to less than 0.06% at 1 kHz, and is less than 1% across the
audio band of 20 Hz to 20 kHz. For 1O-kO loads, the THD+N performance is 0.01 % at 1 kHz, and less than 0.02%
across the audio band of 20 Hz to 20 kHz.

functional block diagram

VDD

8
VDD

Short-Circuit
Protection
CI

RI

LIN--1
CI

RI

LIN+ -1

2

IN1-

3

IN1+

Cc

RO

-=ci

RI

RIN--1
CI

6

IN2-

5

IN2+

V02

RO

RI

-=Over-Temperature
Protection

-=-

To Headphone
Jack
(See TPA152)

Cc

7

RIN+-1

~

T-:
-=-

4

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAD is a trademark of Texas Instruments Incorporated.

-!!1
TEXAS
INSTRUMENTS
POST OFFICE eox 655303 • OAUAS. TEXAS 75265

Copyright © 2000. Texas Instruments Incorporated

&-39

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

SMALL OUTLINEt
(D)

MSOpt
(DGN)

TPA112D

TPA112DGN

-40°C to 85°C
tThe 0 and DGN package
TPA112DGNR).

IS

MSOP
Symbolization
TIAAD

available In left-ended tape and reel only (e.g., TPA112DR,

Terminal Functions
TERMINAL
NAME

1/0

NO.

DESCRIPTION

GND

4

I

GND is the ground connection.

IN1-

2

I

IN1- is the inverting input for channel 1.

IN1+

3

I

IN1 + is the non inverting input for channell.

IN2-

6

I

IN2- is the inverting input for channel 2.

IN2+

5

I

IN2+ is the non inverting input for channel 2.

VDD

8

I

VDD is the supply voHage terminal.

V01

1

0

V01 is the audio output for channell.

V02

7

0

V02 is the audio output for channel 2.

absolute maximum ratings over operating free-air temperature (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Differential input voltage, VI ................................................. -0.3 V to Voo + 0.3 V
Input current, II .......................................................................... ±2.S!lA
Output current, 10 ...................................................................... ±250 rnA
Continuous total power dissipation ................................................ internally limited
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t

Stresses beyond those listed under "absolute maximum ratings" may cause pelTllanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions' is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

TA::;25°C
POWER RATING

=

=

DERATING FACTOR
ABOVE TA 25°C

TA 70°C
POWER RATING

TA 85°C
POWER RATING

=

0

725mW

5.8mW/oC

464mW

377mW

DGN

2.14w*

17.1 mW/oC

1.37W

1.11 W

t Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more infolTllation on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VDD
Operating free-air temperature, TA

~TEXAS

INSTRUMENTS
3-40

POST OFFICE

eox 655303 •

DALLAS, TEXAS 75265

MIN

MAX

2.5

5.5

V

-40

85

°C

UNIT

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

dc electrical characteristics at TA = 25°C, Voo = 3.3 V
PARAMETER
VIO

Input offset voltage

PSRR

Power supply rejection ratio

IDOCa)

Supply current

TEST CONDITIONS

MIN

TYP

MAX

mV

1.5

3

mA

50

dB

83

VOD = 3.2 V to 3.4 V

UNIT

5

IDD(SD)

Supply current in SHUTDOWN mode

10

ZI

Input impedance

>1

IlA
MO

ac operating characteristics, Voo = 3.3 V, TA = 25°C, RL = 8 n
TEST CONDmONS

PARAMETER

MIN

TYP

Po

Output power (each channel)

THDS;0.1%

THD+N

Total harmonic distortion + noise

Po=70mW,

20-20 kHz

2%

THO <5%

>20

BOM

Maximum output power BW

G = 10,

Phase margin

Open loop

MAX

70t

UNIT
mW
kHz

58°

Supply ripple rejection

f= 1 kHz

68

ChanneVchannel output separation

f = 1 kHz

86

dB

SNR

Signal-ta-noise ratio

PO= 100 mW

100

dB

Vn

Noise output voltage

9.5

I1V(rms)

SVRR

dB

t Measured at 1 kHz

dc electrical characteristics at TA = 25°C, Voo = 5 V
PARAMETER
VIO

Input offset voltage

PSRR

Power supply rejection ratio

IDDCa)

Supply current

IDD(SD)
ZI

TEST CONDmONS

MIN

TYP

MAX

UNIT

5

mV

1.5

3

rnA

Supply current in SHUTDOWN mode

60

100

Input impedance

>1

76

VOD = 4.9 Vto 5.1 V

dB

IlA
MO

ac operating characteristics, Voo = 5 V, TA = 25°C, RL = 8 n
PARAMETER

TEST CONDITIONS

MIN

TYP

Po

Output power (each channel)

THO S; 0.1%

THD+N

Total harmonic distortion + noise

PO=150mW,

20-20 kHz

2%

BOM

Maximum output power BW

G=10,

THO <5%

>20

Phase margin

Open loop

70t

MAX

UNIT
mW

kHz

56°

Supply ripple rejection

f= 1 kHz

68

dB

ChanneVchannel output separation

f= 1 kHz

86

dB

SNR

Signal-to-noise ratio

PO=150mW

Vn

Noise output voltage

SVRR

100

dB

9.5

I1V(rms)

t Measured at 1 kHz

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-41

TPA112
150-mW STEREO AUDIO POWEfI AMPLIFIER
SLOS212C - AUGUST 199B - REVISED MARCH 2000

ac operating characteristics, Voo

=3.3 V, TA =25°C, RL =32 Q

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

40t

Po

Output power (each channel)

THD:S; 0.1%

THD+N

Total harmonic distortion + noise

Po =30 mW,

20-20 kHz

BaM

Maximum output power BW

G=10,

THD<2%

Phase margin

Open loop

UNIT
mW

0.5%
kHz

>20
58°

Supply ripple rejection

f = 1 kHz

68

Channel/channel output separation

f=lkHz

86

dB

SNR

Signal-to-noise ratio

PO=100mW

100

dB

Vn

Noise output voltage

9.5

I1V(rms)

SVRR

dB

t Measured at 1 kHz

ac operating characteristics, Voo

=5 V, TA =25°C, RL =32 Q

PARAMETER

THD:s;O.l%

THD+N

Total harmonic distortion + noise

PO=60mW,

20-20 kHz
THD<2%

Maximum output power BW

G = 10,

Phase margin

Open loop

MIN

TYP

40t

Output power (each channel)

BaM

MAX

UNIT
mW

0.4%
>20

kHz

56°

Supply ripple rejection

f= 1 kHz

68

dB

Channel/channel output separation

f= 1 kHz

86

dB

SNR

Signal-to-noise ratio

PO= 150mW

Vn

Noise output voltage

SVRR

t

TEST CONDmONS

Po

Measured at 1 kHz

~TEXAS

3-42

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

100

dB

9.5

I1V (nns)

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
THD+N

Total harmonic distortion plus noise

vs Frequency
vs Power output

1, 2, 4, 5, 7, 8,
10,11,13,14,
16,17,34,36
3,6,9,
12,15, 18

PSSR

Power supply rejection ratio

vs Frequency

19,20

Vn

Output noise voltage

vs Frequency

21,22

Crosstalk

vs Frequency

Mute attenuation

vs Frequency

23-26,
37,38
27,28

Open-loop gain

vs Frequency

29,30

Phase margin

vs Frequency

29,30

Phase

vs Frequency

39-44

Output power

vs Load resistance

31,32

ICC

Supply current

vs Supply voltage

33

SNR

Signal-to-noise ratio

vs Voltage gain

Closed-loop gain

vs Frequency

39-44

Power dissipation/amplifier

vs Output power

45,46

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. 1BCAS 75265

35

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACtERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

10

'#.

VOO=3.3V
PO=30mW
CB=1IL F
RL=320

I

81
Ci
z

+

'#.

c

AV.::5

0

~0

-

Il~I~10

V.

0.1

I

If

Z

+
c

~

,

~'
is

"

.2

5

PO=15mW '

==

Po=10mW

::

!

AV=1

0.01

0.1

i

"-

iii

'0

.~

.......-::

i!

~

VOO=3.3V
AV=1 VN
RL=320
CB=1IL F

I

~I

z

+

Z

~

::

1&

"III

0.01

+

Q

: PO=30mW

Q

I-

0.001
20

100

1k

10k 20k

IIII

III

0.001
20

1k

100

f - Frequency - Hz

Figure 1

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

10

10

'#.

'#.

VOO = 3.3 V
RL=320
AV=1 VN
CB=1ILF

I

~

Z
+

c
0

I
+

27 kHZ

i

.~

.2

I

0

~

0.1

1 kHz

j

-

I

Z
+
Q

~
0.01

120HZ

.L

.......

1

!

~

J

-

10

AV=10mW

11
0.1 E

AV=5mW

.

NI

~
~
50

L..oo'

I..Y ~

L

1/

0.01
AV=1 mW

II

0.001
20

Po - Output Power - mW

I I I II

100

1k
f - Frequency - Hz

Figure 4

Figure 3

~TEXAS

3-44

L

c

~

01

VOO=5V
Po=60mW
RL=320
CB= 1ILF

I

10kHz

1:
0

::

10k 20k

f - Frequency - Hz

INSTRUMENTS
POST OFACE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER
10

10

'i!.

~VOO=5V

VOO=5V
RL=32r.!
AV=1 VN
CB=1I1F

I

~

Z

+
c

~AV=1 VN
t- RL=32r.!
t- CB =1I1F
20,kHZ

0

t:

~

.2
c

10kHz

19?"

C

Po=30mW

0.1 1=

I--I-

0

i

I§.. PO=15mW
Wll

J:

!

~I

I'

0.01

~

Z

+

II

~

0.1

~

1 kHz

~

Po=60mW

Q

J:

II

I-

0.001
20

20Hz

t--

11

100

1k

Po - Output Power - W

Figure 5

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

VOO=3.3V
10~;W_
RL=10kr.!

VOO=3.3V
RL=10kQ
Po = 100I1F
CB=1I1F

I

J0
z

t--J I-

0.01 L---L---L-.LJ.....J...LLL-_-l---l.--l.-l-.J...J..J..LJ.._--'
0.002
0.01
0.1
0.2

10k 20k

f - Frequency - Hz

'i!.

-

1/

AV=1 VN
CB= 111F

+

c
0

t:
0

~
~0

AV=5mW
0.1

i

J:

!

~

V

0.01

I

Z

+

AV=2mW

Q

J:

-

II 1111

I-

0.001
20

100

1k

10k 20k

0.001

L....I-l...u.J.I.LL...---I---L...J....I...LJJJJ....---L-I....1-1..u.J..Ll---'

20

f - Frequency - Hz

100

1k

10k 20k

f - Frequency - Hz

Figure 7

Figure 8

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-45

TPA112
150-mWSTEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10

vs

OUTPUT POWER

FREQUENCY
10

Voo = 3.3 V
RL=10kQ
AV=1 VN
CB=1 !IF

'#.
I

GI

~

'#.

z

+
c

+
c

~

~

~

.s
is

0.1
20Hz _

0

i

iii

;2I

VOO=5V
RL=10kn
PO=3oo!lW
CB=1 !IF

I

.;0

Z

.2
c

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

10kHz

i""_

.~

=I

0.1

AV=5

0

i

AV=1

:c
pJ

0.01

!

~

""

0.01

LIt. I-"'" ~

I

Z

Z

20Hz

+

CI

+

CI

1 kHz

j!:

I

0.001

I-

I I

100

10

5

AV=2

:c

200

I I I

0.001
100

20

Po - Output Power -!lW

f - Frequency - Hz

Figure 10

Figure 9
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

VOO=5V
10~~_
RL=10kn
Av=1 VN
CB=1 !IF

10

'#.

VOO=5V
RL=10kn
Av=1 VN
CB=1 !IF

I

I

+

i..
is
C

0.1

o

i!

~

0.01

100

1k

10k 20k

j....I

;;f

10kHz 1 k~Z- r--

j!:

20

1\

\

~

L...J-..I..J...U..LLI----J'--'-1...Ju..J,;LJ..I..----L...................l..U.......--O

--

20Hz
20kHz

I
Z

0.001

10k 20k

1k

II

0.001

5

100

10

f - Frequency - Hz

Po - Output Power -I1W

Figure 12

Figure 11

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655003 • DALLAS. lEXAS 75265

I 1-r-500

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS
TOTAL HARMONIC DISTORTION PLUS NOISE

#.

2

I

§

CI)

.!!!
0

z
III

::3

ii:
c
0

,

.2
c

-

AV=

""

/

"

/~

AV=1

["0......

I'

0

!

- r-

0.1

is

~

10

11111

AV=

'Iii

J:

FREQUENCY

./

0

i:

vs

FREQUENCY

~O~~I~.~I~

Po=75mW
~ RL=8Q
=CB=1I1F

I
CI)

III

'0

z

Po=30mW

+

"

~
,- PO=15mW

j

;§

~

0.01

I

z+

Z
+

PO=75mW

C
J:

I-

0.001
20

100

1k
f - Frequency - Hz

II

0.001

10k 20k

20

100

vs

OUTPUT POWER

FREQUENCY
I

t-RL=8Q
~AV=1 VN

+

c

-'

i:

10kHz .~

s

!
I

..

r=t-

T

11111111

1111 AJ=

Voo =5 V
Po=100mW
RL=8Q
CB=1I1F

rr-

c.! v=5

"J

A ~""

0.1

'Av= 1

oS!

'c0

1 kHz

E
0.1

,....

"""

I

~

JI

I-

I-

0.01
10m

~

0.01

~

20Hz

I

r----

.L~

!

;§
Z
+
J:

F

£

ic

C

I=:

I

r•

~~kHz

t--

0

2

#.

1= voo = 3.3 V

3l
'0
z

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10
I

10k 20k

Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

#.

1111
1k

f - Frequency - Hz

Figure 13

&II

..-!l~

0.1

'iii

I

J:
'iii

~

c

~0

~

C
J:
I-

VOO=3.3V
RL=8Q
AV=1 VN

#.

~

r----

. /~

0.01

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

0.1

0.001

0.3

20

100

1k

10k 20k

f - Frequency - Hz

Po - Output Power - W

Figure 15

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • OALLAS. TEXAS 75265

3-47

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C

AUGUST 1998

REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONI~ DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

POWER OUTPUT

10
;P.

10

~VOO=5V
RL=8kn
I- AV= 1 VN

;P.

t:

I

.!z
+

.!z

PO=30mW

c

+

c0

0

'E
0

~u

'c0
01

'c0

r---

~

..,..

...

:c

01

:c

+

PO=10mW

:c

j:

111111

I-

0.001
20

100

1k

20Hz

0.01
10m

10k 20k

0.1

f - Frequency - Hz

Po - Output Power - W

Figure 17

Figure 18

POWER SUPPLY REJECTION RATIO

POWER SUPPLY REJECTION RATIO

vs

vs

FREQUENCY

FREQUENCY

0
III
"I:J

-10

I

i

-20

c

-30

II:
0

:g
CD

-40

i:'
Q.

-50

l

Q.

:s

I/)

;

0
Do.
I

II:
II:
I/)

Do.

-60
-70

VOO=3.3V
RL = 8 Q to 10 kQ

'""""r--

I I

f"-....
~r--

l'

r-

-20

c

-30

0

u

"'1\

~
~
\}

~~

~

V

l
t

1k
f - Frequency - Hz

I.:::t'-

-80

I

-70

II:
II:

B=

I'~,...

IJ.

"

'"

~

~

~~
1\

iiuilli I

-90
-100
20

i'
k

-4J0

2.5

,
100

1k
f - Frequency - Hz

Figure 20

Figure 19

-!!1
TEXAS
INSTRUMENTS
3-48

I
J
1~1I=1IJ.F

B= .11J.F

~

-50

:s

I/)

Do.
I

10k 20k

-40

.........

Q.

If
100

I"j'

i

II:

VOO=5V
RL=SOt010kQ

-10

:;:0

t'-....

I't-I B= IJ.
~I"'"
Jy~Js~lll~.65

-100
20

"I:J

CD

-80

-90

.! I

~

0
III
I

B= .11J.F
I III
I'l!:
=11J.F

~

t-t-

~

'::1

"""

Q

)kHZ

I I

r---

I

Z

Q

10kHz

0.1

~

0.01

Z

+

-

~u

k:::--

0.1 1= PO=60mW

I k~kHz

r--

'E
0

~

1§
~
I

VOO=5V
RL=8Q
AV=1 VN

I

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

OUTPUT NOISE VOLTAGE

vs

vs

FREQUENCY

FREQUENCY

20

~
I

20

10

~

III
DI

10

I

III

!

~

I

iz

Jl

~

~

'S
a.
'S
0

:;

a.
'S
0

I

::f'

I

t- VOO=3.3V
BW = 10 Hz to 22 kHz
AV=1 VN
RL=80to 10 kn
1
20

100

VOO=5V
BW = 10 Hz to 22 kHz
RL = 8 0 to 10 kn
AV=1 VN

>c

1k

1 ~"uillii
20
100

10k 20k

f - Frequency - Hz

I

I I II

1k

Figure 21

Figure 22

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-60

-50

PO=25mW
VOO=3.3V
RL=320
-70 t-CB=1f,1F
AV=1 VN
-75

-65

m

"...
I

~

..e

u

-80

-55 -

-90
-95

VOO=3.3V
RL=80
- CB=1 f,lF
-65 :--, Av= 1 VN

m

,

V

N2 00

~

LI.-IN ,1

-105

I

or

"""

1k

-70
-75

u

10k 20k

~~

-as

'-

IN2TOOUT

~

-80

1V
~

fOiiii'"

-90

i,

I I
100

"...I

Ie

1/

-100

-110
20

'p~'~~'~m~

-60

t'\

-85

10k 20k

f - Frequency - Hz

IN1TOOUT2

-95
-100

20

f - Frequency - Hz

Figure 23

100

1k
f - Frequency - Hz

Figure 24

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-60

-50

VOO=5V
PO=25mW
CB=1I1F
RL=320
AV=1 VN

-65

--65
-75
III
'0

...
i
I

e

0

-

-65

1,1'01.

--60

V

I'

-65

I--

IN2TOOUT1
\

-90

~~

-95

VOO=5V III
.oS5 - PO= 100mW
CB= 111F
-60 f-RL=SO
AV=1 VN

-

~

~

III
'0
I

-70

...

If

r-.

--60

0

IN2TO~UT1

l'

-65

~ ~~

-100

..... 1'

-75

-90

IN1TOOUT2
-105
-110
20

II
100

-95

IIII

1k

-100
20

10k 20k

f -Frequency - Hz

+--

IN111~

fllul -

IIIIll!
100

1k

10k 20k

Figure 26

MUTE ATTENUATION

MUTE ATTENUATION

vs

vs

FREQUENCY·

~

t...-'"

f - Frequency - Hz

Figure 25

~
I

17

FREQUENCY

~-H+H~~~~~--~~~~~

~

I :: ~-H+H~~~~~--~~~~~

ti

j

!

c

C

-70 H-HI++I+l-----+-+-1I-+++Itt---H-+t+tftl--l

~o~+-~H*~-l-++~ffi-_r~~~-i

I

~o~+-~H*~-l-++~ffi-_r~~~-i

-50

I----:.+-I-+-H*~-l-++~ffi-_r~~~-i

--60~H#~=++~fIjj::+++tt#H
-701-+-l-+++H+I-H+t1I-tttt---t-+ttttttl--l

--60 ~~~*-~4-~~--~-H~~~

-90

-100 \-...1.....L.J..J..J.J.JJ..-....J-...L....JU-.I..L.I.I.I._............................I.I...-:-:'
20
100
1k
10k 20k

-100 L.-..L....L...J..Ju.u.u......-L-..L....J..J..U.~--'-...J....I..J....I.JW.I--'
20
100
1k
10k 20k

f - Frequency - Hz

f - Frequency - Hz

Figure 27

Figure 28

~TEXAS

3-50

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE MARGIN

vs
FREQUENCY
100

80

\",

~

ID
'0
I

=

TA 25°C
No Load

Phase

C

'OJ

40

0
0

....t

Gain

C
CD

a.
0

20

90°

1\

'I"-

CI

120°

i i ii

I'-

60

a.

150°

V~~I~I~~~ J I

c

~

::E
60°

~

CD
01

II>

&.
Do.

I

30°

....E

I'0

0°

-20
100

lk

lOOk

10k

-30°
10M

1M

f - Frequency - Hz

Figure 29
OPEN-LOOP GAIN AND PHASE MARGIN

vs
FREQUENCY
100

~~~~II~VI

I'80
ID

1\

'0

I

c

60

'OJ
0
0

I I I II

'I"-

Phase

~,

CI

a.

I

TA=25°c
No Load

40

~CD

Gain

a.
0 20

I'-

'r\
,

0

-20
100

-300

lk

10k

lOOk

1M

10M

f - Frequency - Hz

Figure 30

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

3-51

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

vs

LOAD RESISTANCE

LOAD RESISTANCE

120

100

~I

I

II.

S
I

-

250

40

"'~

2

~

~

200

I

II.

150

I

100

THD~N=1 ~

II

VDD=5V
Av=1 VN

~

~

"-

!

0

VDD=3.3V
AV=1 VN

\

60

300

THD~N=1 ~

1\

80

OUTPUT POWER

vs

\
~

J

" '"
i'-..

2

~ to--.

20

50

0

24
32
48
40
RL - Load Resistance - Q

16

8

56

o

64

8

16

24

56

64

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

SUPPLY VOLTAGE

FREQUENCY

'i!-

t=

I

~

VI = 1 V
AV= 1 VN
- RL=10kQ

I::

~=

1.2

_CB=1~F

~

cc

II.

E

C

~

I

0.1

I

0.8

0

I

0.6

_

I/)

I

48

vs
1.4

Q
Q

40

Figure 32

SUPPLY CURRENT

~
CL
CL
::I

32

\'-. r--

RL - Load Resistance - Q

Figure 31

'E
~::I

-

0.4

1

0.01

!

{!!.
I

0.2
0

i--"

~

:c
I-

2.5

3

3.5

4

4.5

5

5.5

0.001
20

1k
f - Frequency - Hz

VDD - Supply Voltage - V

Figure 34

Figure 33

-!II
TEXAS
INSTRUMENTS
3-52

100

POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

10k 20k

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS
SIGNAL-TO-NOISE RATIO
104

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

VOLTAGE GAIN

FREQUENCY

?l-

I

~b
i

'OJ

Z

til

::s

~

is:
c

c

i!
0

;;
is

~

i'...

96

zfJl

94

'",

_

l'-..

92
1

2

0.1

0

'\

""

98

~
I
II:

~

'0

\

100

Voo=5V
AV=1
r- RL=10kCl
r- CB=1I!F

3l

102 I\.
III
'a
I

1=

I

VI=1 V

3

4
5
6
7
8
AV - Voltage Gain - VN

0.01

J

li
~

'" i'.
9

I

Z

~I-

0.001

10

mlrD"~
L-....L...J...I.........JJJ-.........--1-...........w.L._..................u.u."----'

20

100

Figure 35

III
'a
I

...

~til

e

-70

r-

-a0

r-

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY
-60

to~ ~ IJ.~I ~
VO=1 V
RL=10kCl
CB=1I!F

~

........

-110

1',....

()

-120

,~

Ii

IN2to OUT1
.1
f--I IV

~~

til

'"

e

20

1111

-90
-100
-110

"'r--

()

~,..

-130

100

"'I"

~

1111
I"IN2 )0 OUT1

~ I"'f'

i;'~

P'"

1k
f - Frequency - Hz

10k 20k

lJV
IJ

IN1 toOUT2

-140

I

IIIIII

-150

III
'a
I

i

IN1~OUT2

-140

11111111

-120
--'!.

-130

!

VoO=5V
-70 t - VO=1 V
RL=10kn
-60 t - CB=1I!F

-SO
-100

10k 20k

Figure 36

CROSSTALK

-60

1k
f - Frequency - Hz

-150
20

Figure 37

11111
100

I III
1k
f - Frequency - Hz

10k 20k

Figure 38

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-53

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

200°

1111111
Phase

1/

180°
160°

1\

!

140° .c
~

120°

IJ

VOO=3.3V
RI=20kQ
RF=20kQ
RL=32Q
CI=lI1F
AV=-l VN

30
20

100°
80°

Jl!llil

10

o

V-

-10

10

111111
100

lk

10k

lOOk

1M

f - Frequency - Hz

Figure 39
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

200°

111111111
Phasa

1/

180°
160°

I'

140°
120°

II

VOO=5V
RI=20kQ
RF=20kQ
RL=32Q
CI= ll1F
AV=-l VN

ID

'CI

I

30

~

20

c

a.

i

o

llil

V
10

100

11I111111
lOOk
10k
f - Frequency - Hz
lk

Figure 40

~TEXAS

3-54

80°

111111111
Gain

10

-10

100°

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

1M

3!
!~

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

200°
180°

II "'"

~

Phase

160°

1\

140°
120°

Voo = 3.3 V
RI=20kO
RF=2OkO
RL=80
CI=1 J.lF
AV=-1 VN

I

III

"a

I

c
~

r
40

J

1000
80°
60°

I~~~~II

f~

11111111
~

O

10

100

1k

II "'10k
"

100k

1M

f - Frequency - Hz

Figure 41
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

.....

200°
180°

I """

.....

Phase

./

1600
140°
120°

:::

III
"a

I

30

~

20

f

10

c

(j

Voo = 3.3 V
RI=20kO
RF=2OkO
RL=10kO
CI=1 J.lF
AV=-1 VN

SOO

1111111

o
-10

100°

Gain

""'"

10

100

1k

I """10k

100k

f - Frequency - Hz

Figure 42

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

1M

J

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

200°

11111111

/~

r...

Phase

11111111

I

!XI
'tI
I
C

120°

80°
60°
40°

'ci!l~"

f~

:II

01
.c
a.

100°

11111111

~

160°
140°

11111111

VOO=5V
RI=20kO
RF=2OkO
RL=80
CI=1IlF
AV=-1 VN

j

180°

11111111

I,..-

O

10

11111111
100

1k

10k

100k

1M

f - Frequency - Hz

Figure 43
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

i"'"

1/

!XI

Phase

rl'

11111111

J~~~lt

'tI

I

c

~

20

RI=2OkO
RF=20kO
RL=10kO
CI=1IlF
AV=-1 VN
11111111

10

11111111

a.

i

200°

lilll

~

30

160°
140°
120°
100°
80°

Gain

o

l111L

11111111

-10
10

100

1k
10k
100k
f - Frequency - Hz

Figure 44

-!I TEXAS

3-56

180°

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

1M

:I

.c
a.

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION/AMPLIFIER
vs
OUTPUT POWER

ao

1~

=

Voo 3.3 V _
70

:=E
I

I
0

a..

~'ii
E

160

r-....

/ ,-

40

/

30

i'o..

'/

CC

20

?

~a

o

"

i\ '" ~

~

40

~

I

"-

.

~

E

60

cc

1'\

/

100

~

~

~1001~1401~1~

~

-...... r-

/V
V
V '-

20

o

16a

I

40

~

~i"""

/

120

I

'ii

i4a ~

10

:=E

t'\.

~a

,/

140

""

/

50

o

~

ah

VOO=5V

ala

/

60

POWER DISSIPATION/AMPLIFIER
vs
OUTPUT POWER

o

~

~

40

Load Power - mW

-

~

r-..

a

~ r-.....

...........

--

t-.....

~1001~1401~1~

~O

Load Power - mW

Figure 45

Figure 46

APPLICATION INFORMATION

gain setting resistors, RF and RI
The gain for the TPA 112 is set by resistors RF and RI according to equation 1.
Gain

= -

(~~)

(1)

Given that the TPA112 is a MOS amplifier, the input impedance is very high. Consequently input leakage
currents are not generally a concern, although noise in the circuit increases as the value of RF increases. In
addition, a certain range of RF values is required for proper start-up operation of the amplifier. Taken together
it is recommended that the effective impedance seen by the inverting node of the amplifier be set between
5 kll and 20 kQ. The effective impedance is calculated in equation 2.

R R
Effective Impedance = R F ~
F+ I

(2)

As an example, consider an input resistance of 20 kQ and a feedback resistor of 20 kQ. The gain of the amplifier
would be -1 and the effective impedance at the inverting terminal would be 10 kll, which is within the
recommended range.

-!/} TEXAS

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3-57

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
gain setting resistors, RF and R, (continued)
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kil, the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF- This, in effect, creates a
low-pass filter network with the cutoff frequency defined in equation 3.
f

1
co(lowpass) - 2:n:R FC F

(3)

For example, if RF is 100 kil and CF is 5 pF then fco(lowpass) is 318 kHz, which is well outside the audio range.

input capacitor, C,
In the typical application, an input capacitor, Cj, is required to allow the amplifier to bias the input signal to the
proper dc level for optiinum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 4.
f

1
co(highpass) - 2:n:R I C I

(4)

The value of CI is important to consider, as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 20 kg and the specification calls for a flat bass response down to 20 Hz.
Equation 4 is reconfigured as equation 5.
CI =

1
2:n:RI fCO(highpass)

(5)

In this example, CI is 0.40 IlF, so one would likely choose a value in the range of 0.47 IlF to 1 IlF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI' CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier
that reduces useful headroom, especially in high-gain applications (> 10). For this reason a low-leakage
tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the
capacitor should face the amplifier input in most applications, as the dc level there is held at Vool2, which is
likely higher that the source dc level. It is important to confirm the capacitor polarity in the application.

power supply decoupling, Cs
The TPA 112 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure that the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 IlF, placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 IlF or greater placed near
the power amplifier is recommended.

"'TEXAS

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TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

APPLiCATiON iNFORiviATION

midrail voltage
The TPA112 is a single-supply amplifier, so it must be properly biased to accommodate audio signals. Normally,
the amplifier is biased at Vool2, but it can actually be biased at any voltage between Voo and ground. However,
biasing the amplifier at a point other than Vool2 will reduce the amplifier's maximum output swing. In some
applications where the circuitry driving the TPA112 has a different mid rail voltage, it might make sense to use
the same midrail voltage for the TPA112, and possibly eliminate the use of the dc-blocking caps.
There are two concerns with the midrail voltage source: the amount of noise present, and its output impedance.
Any noise present on the midrail voltage source that is not present on the audio input signal will be input to the
amplifier, and passed to the output (and increased by the gain of the circuit). Common-mode noise will be
cancelled out by the differential configuration of the circuit.
The output impedance of the circuit used to generate the midrail voltage needs to be low enough so as not to
be influenced by the audio signal path. A common method of generating the midrail voltage is to form a voltage
divider from the supply to ground, with a bypass capacitor from the common node to ground. This capacitor
improves the PSRR of the circuit. However, this circuit has a limited range of output impedances, so to achieve
very low output impedances, the voltage generated by the voltage divider is fed into a unity-gain amplifier to
lower the output impedance of the circuit.

voo

voo
R

R

TLV2460

Mldrall
Mldrall
CSYPASS

T

R

CSYPASS

a) Midrail Voltage Generator Using a Simple
Resistor-Oivider

T

R

b) Buffered Midrail Voltage Generator to Provide
Low Output Impedance

Figure 47. Midrall Voltage Generator

If a voltage step is applied to a speaker, it will pop. To reduce popping, the midrail voltage should rise at a
sub-sonic rate; that is, a rate less than the rise time of a 20-Hz waveform. If the voltage rises faster than that,
there is the possibility of a pop from the speaker.
Pop can also be heard in the speaker if the mid rail voltage rises faster than either the input coupling capacitor,
or the output coupling capacitor. If midrail rises first, then the charging of the input and output capacitors will
be heard in the speaker. To keep this noise as low as possible, the relationship shown in equation 6 should be
maintained.
(6)

Where CBYPASS is the value of the bypass capacitor, and RSOURCE is the equivalent source impedance of the
voltage divider (the parallel combination of the two resistors). For example, if the voltage divider is constructed
using two 20-1<0 reSistors, then RSOURCE is 10 1<0.

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3-59

TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
midrail bypass capacitor, CB
The midrail bypass capacitor, Ca, serves several important functions. During start-up, Ca determines the rate
at which the amplifier starts up. This helps to push the start-up pop noise into the subaudible range (so slow
it can not be heard). The second function is to reduce noise produced by the power supply caused by coupling
into the output drive signal. This noise is from the mid rail generation circuit internal to the amplifier. The capacitor
is fed from the resistor divider with equivalent resistance of RSOURCE. To keep the start-up pop as low as
possible, the relationship shown in equation 7 should be maintained.
1

<_1_

(7)

(C a x RSOURCE) - (C ,R,)

As an example, consider a circuit where Ca is 1 IlF, RSOURCE = 160 kQ, C, is 1 IlF, and R, is 20 kQ. Inserting
these values into the equation 9 results in:
6.25 s 50
which satisfies the rule. Bypass capacitor, Ca, values of 0.1 IlF to 11lF ceramic or tantalum low-ESR capacitors
are recommended for the best THO and noise performance.

output coupling capacitor, Cc
In the typical single-supply single-ended (SE) configuration, an output coupling capacitor (Cc) is required to
block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling
capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by
equation 8.
f

-

(out high) -

1

23tR LCc

(8)

The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the
low-frequency comer higher. Large values of Cc are required to pass low frequencies into the load. Consider
the example where a Cc of 68 IlF is chosen and loads vary from 32 0 to 47,kO. Table 1 summarizes the
frequency response characteristics of each configuration.

Table 1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode
RL

Cc

Lowest Frequency

320

681lF

73 Hz

10,0000

681lF

0.23 Hz

47,0000

681lF

0.05 Hz

As Table 1 indicates, headphone response is adequate and drive into line level inputs (a home stereo for
example) is very good.

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TPA112
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS212C - AUGUST 1998 - REVISED MARCH 2000

AFPLiCATiON iNFORiviAiiOi,j
output coupling capacitor, Cc (continued)
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following
relationship:

output pull-down resistor, Rc + Ro
Placing a 100-n resistor, Re, from the output side of the coupling capacitor to ground insures the coupling
capacitor, Ce, is charged before a plug is inserted into the jack. Without this resistm, the coupling capacitor
would charge rapidly upon insertion of a plug, leading to an audible pop in the headphones.
Placing a 20-kn resistor, Ro, from the output of the IC to ground insures that the coupling capacitor fully
discharges at power down. Ifthe supply is rapidly cycled withoutthis capacitor, a small pop may be audible in
10-kn loads.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this application. A real capacitor can be modeled simply
as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial
effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real
capacitor behaves like an ideal capacitor.

s-V versus 3.3-V operation
The TPA112 was designed for operation over a supply range of 2.7 V to 5.5 V. This data sheet provides full
specifications for 5-V and 3.3-V operation since these are considered to be the two most common standard
voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain
setting, or stability. Supply current is slightly reduced from 3.5 mA (typical) to 2.5 mA (typical). The most
important consideration is that of output power. Each amplifier in the TPA 112 can produce a maximum voltage
swing of Voo -1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as opposed
when Vo(PP) 4 V while operating at 5 V. The reduced voltage swing subsequently reduces maximum output
power into the load before distortion begins to become significant.

=

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:H31

3--62

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
• 150 mW Stereo output
• PC Power Supply Compatible
- Fully Specified for 3.3 V and 5 V
Operation
- Operation to 2.5 V

V01
INBYPASS
GND

•
•
•
•

Pop Reduction Circuitry
Internal Mid-Rail Generation
Thermal and Short-Circuit Protection
Surface-Mount Packaging
- PowerPADTM MSOP
- SOIC
• Pin Compatible With LM4880 and LM4881
(SOIC)

VDD

V02
IN2SHUTDOWN

description
The TPA 122 is a stereo audio power amplifier packaged in either an 8-pin SOIC, or an 8-pin PowerPADTM MSOP
package capable of delivering 150 mW of continuous RMS power per channel into 8-0 loads. Amplifier gain
is externally configured by means of two resistors per input channel and does not require external compensation
for settings of 1 to 10.
THD+N when driving an 8-0 load from 5 V is 0.1 % at 1 kHz, and less than 2% across the audio band of 20 Hz
to 20 kHz. For 32-0 loads, the THD+N is reduced to less than 0.06% at 1 kHz, and is less than 1% across the
audio band of 20 Hz to 20 kHz. For 1O-kO loads, the THD+N performance is 0.01 % at 1 kHz, and less than 0.02%
across the audio band of 20 Hz to 20 kHz.

typical application circuit

320kn

RF

l-

Audio
Input

~

l~

RI

2

-AA

320kn

Vo0f2

IN1-

Audio
Input

1-

3

BYPASS

1

6

IN2-

i

VOO

c

J,

~C

CBt

~

L

RI

-=-

I
CI

V02 7

I+

From Shutd own
Control Cire ult

5

I

SHUTDOWN

I

RF

..

V01

+

CI

~

VDO 8

L

I~
Bias
Control

I n4

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS .
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Copyright @ 2000, Texas Instruments Incorporated

3-63

TPA122
150-mW STEREO
AUDIO POWER AMPLIFIER
,I
SLOS211C - AUGUST1998 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

SMALL OUTLlNEt
(D)

MSOpt
(DGN)

-40°C to 85°C

TPA122D

TPAI22DGN

MSOP
Symbolization
TIME

tThe D and DGN package IS available In left-ended tape and reel only (e.g., TPAI22DR,
TPA 122DGNR).

Terminal Functions
TERMINAL
NAME
BYPASS

110

NO.

DESCRIPTION

3

I

Tap to voltage divider for intemal mid-supply bias supply. Connect to a 0.1
best performance.

GND

4

I

GND is the ground connection.

IN1-

2

I

IN1- is the inverting input for channell.

IN2-

6

I

IN2- is the inverting input for channel 2.

SHUTDOWN

5

I

Puts the device in a low quiescent current mode when held high

VDD

8

I

VDD is the supply voltage terminal.

VOl

1

0

Vo 1 is the audio output for channell.

V02

7

0

V02 is the audio output for channel 2.

~F

to 1 ~F low ESR capacitor for

absolute maximum ratings over operating free-air temperature (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ........................................................... -0.3 V to Voo + 0.3 V
Continuous total power dissipation ................................................ internally limited
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg ................................................... -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ................. . . . . . . . . . . . . .. 260°C
t Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

TA :5 25°C
POWER RATING

DERATING FACTOR
ABOVE TA 25°C

TA = 70°C
POWER RATING

TA = 85°C
POWER RATING

=

D

725mW

5.8mW/OC

464mW

377mW

DGN

2.14w*

17.1 mW/oC

1.37W

1.IIW

:J: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VDD
Operating free-air temperature, TA

~TEXAS

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MIN

MAX

2.5

5.5

V

-40

85

°C

UNIT

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211 C - AUGUST1998 - REVISED MARCH 2000

dc electrical characteristics at TA

= 25°C, Voo = 3.3 V

PARAMETER
VIO

Input offset voliage

PSRR

Power supply rejection ratio

IDD

Supply current

IDD(SD)
ZI

TEST CONDITIONS

MIN

TYP

MAX

UNIT

5

mV

1.5

3

mA

Supply current in SHUTDOWN mode

10

50

Input impedance

>1

ac operating characteristics, Voo

dB

83

VDD = 3.2 V to 3.4 V

!LA
MQ

=3.3 V, TA =25°C, RL =8 n

PARAMETER

TEST CONDITIONS

MIN

TYP

Po

Output power (each channel)

THD~O.I%

THD+N

Total harmonic distortion + noise

Po=70mW,

20-20 kHz

2%

THD<5%

>20

MAX

70t

UNIT
mW

Maximum output power BW

G=10,

Phase margin

Open loop

Supply ripple rejection

f= 1 kHz

68

Channel/Channel output separation

f = 1 kHz

86

dB

SNR

Signal-to-noise ratio

PO=100mW

100

dB

Vn

NOise output voltage

9.5

I!V(rms)

BOM

kHz

58°
dB

t Measured at 1 kHz

dc electrical characteristics at TA

=25°C, Voo =5 V

PARAMETER
VIO

Input offset voltage

PSRR

Power supply rejection ratio

IDD

Supply current

IDD(SD)
ZI

TEST CONDITIONS

MIN

TYP

UNIT

5

mV

1.5

3

mA

Supply current in SHUTDOWN mode

60

100

Input impedance

>1

ac operating characteristics, Voo

76

VDD = 4.9 Vto 5.1 V

dB

!LA
MQ

=5 V, TA =25°C, RL =8 n

PARAMETER

TEST CONDITIONS

MIN

TYP

Po

Output power (each channel)

THD~O.I%

70t

THD+N

Total harmonic distortion + noise

PO= 150mW, 20-20 kHz

2%

Maximum output power BW

G= 10,

>20

Phase margin

Open loop

56°

BOM

MAX

THD<5%

MAX

UNIT
mW
kHz
dB

Supply ripple rejection ratio

f= 1 kHz

68

Channel/channel output separation

f = 1 kHz

86

dB

SNR

Signal-to-noise ratio

Po=150mW

100

dB

Vn

Noise output voltage

9.5

I!V(rms)

t Measured at 1 kHz

-!11 TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-65

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S211C - AUGUST1998 - REVISED MARCH 2000

ac operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 32 Q
PARAMETER

TEST cONDmONS

MIN

TVP

Po
THO+N

Output power (each channel)

THOSO.l%

Total harmonic distortion + noise

PO=30mW,

2D-20kHz

0.5%

BOM

Maximum output power BW

G= 10,

THO <2%

>20

Phase margin

Open loop

MAX

40t

UNIT
mW
kHz

58°

Supply ripple rejection

f=l kHz

68

dB

ChanneVchannel output separation

f= 1 kHz

86

dB

SNR

Signal-Ie-noise ratio

PO=lOOmW

Vn

Noise output voltage

100

dB

9.5

I1V(rms)

t Measured at 1 kHz

ac operating characteristics, VDD = 5 V, TA = 25°C, RL = 32 Q
PARAMETER

TEST CONDITIONS

MIN

TVP

Po
THO+N

Output power (each channel)

THOsO.l%

Total harmonic distortion + noise

PO=60mW,

2D-20kHz

0.4%

BaM

Maximum output power BW

G=10,

THO <2%

>20

Phase margin

Open loop

40t

MAX

UNIT
mW
kHz

56°

Supply ripple rejection

f= 1 kHz

68

dB

ChanneVchannel output separation

f= 1 kHz

86

dB

SNR

Signal-to-noise ratio

PO=150mW

Vn

Noise output voltage

t Measured at 1 kHz

~TEXAS

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100

dB

9.5

I1V(rms)

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211 C - AUGUST1998 - REVISED MARCH 2000

TYPiCAL CHARACTERiSTiCS
Table of Graphs
FIGURE

vs Frequency
THO+N

Total harmonic distortion plus noise
vs Power output

Vn

1,2,4, 5, 7, 8,
10,11,13,14,
16,17,34,36
3,6,9,
12, 15, 18

Supply ripple rejection

vs Frequency

19,20

Output noise voltage

vs Frequency

21,22

Crosstalk

vs Frequency

23-26,
37,38
27,28

Mute attenuation

vs Frequency

Open-loop gain and phase margin

vs Frequency

29,30

Output power

vs Load resistance

31,32

Closed-Loop gain and phase

vs Frequency

39-44

Output power

vs Load resistance

31,32

100

Supply current

vs Supply voltage

33

SNR

Signal-to-noise ratio

vs Voltage gain

Closed-loop gain

vs Frequency

39-44

Power dissipation/amplifier

vs Output power

45,46

35

-!!1

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3-67

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S211C

AUGUSTl998- REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY

vs
FREQUENCY

10

10

il-

VOO=3.3V
PO= 30 mW
CB=1I1 F
RL=320

I

.!z

+
c
0

.~

.10
z

L

+
c

lVI~~~~I~N

/.

0.1

~

~

V

"/

Q

~0

0

~

~

!
t=I

AV=-1 VN

0.01

I

Z

PO=15mW

....

I'-

0.01

Z

+

+

PO=30mW

Q

:c

:c
....

0.001
20

1k

100

II

0.001

10k 20k

IIII

100

20

1k

f - Frequency - Hz

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

VB

vs

OUTPUT POWER

FREQUENCY

10

il-

10

il-

.... VOO=3.3V
I-RL=320
.... AV=-1 VN
.... CB=1I1F

I

~

Z

+
c

.1

~

+
c

10kHz

i

5

.2

.~

c
0

0.1

-

+

JDHZ

.L

....

0.01

E

1

!

~

J

-

~

Q

:c

10

0.1 0=

AV=-10VN
1 -I

AV =-5 VN

•--

0.01

./

l."": V
V

I

Z

+

Q

i!:
50

Av= 1 VN

II

0.001
20

Po - Output Power - mW

Figure 3

I I III

100

1k
f - Frequency - Hz

Figure 4

~TEXAS

INSTRUMENTS
3-68

L.

I--

~

:c

1 kHz

Z

~~
l-

~

~

!
t=I

VOO=5V
Po=60mW
RL=320
CB=1I1F

I

.,kHZ

~

Ii

10k 20k

f - Frequency - Hz

Figure 1

:c

~

Po=10mW

:!

"-

Q

....

e=

0.1

E

01

:c

!

VOO = 3.3 V
AV= 1 VN
RL=320
CB=1I1 F

I

AV=-5~N

t!0

~

il-

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

10k 20k

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S211 C - AUGUST1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER
10

10
~

VOO=5V
RL=32n
AV=-1 VN
CS= 1ILF

I

~
Z

!=VOO=5V
Av=-1 VN
t- RL=32n
t- CS =1ILF

1=

+
c

~kHZ

0

t:
0

~

..

C

~

PO=30mW

0.1 F=

~

1==::=

0

..
:c

E

!

{!.

5. Po=15mW
l'IJl1

~

0.01

I

li

~

0.1

Urn. ~~

+

11kHz

1111

II

0.001
20

20Hz

\"'-0"

PO=60mW

I-

I"-

to-

Z
Q

:c

-

10kHz

100

1k

0.01
0.002

10k 20k

0.1

0.2

Po - Output Power - W

Figure 5

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORT10N PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

10~~lmm

VOO=3.3V
RL=10kn
PO=1ooILF
CB=1ILF

3:
'0
z

-

r--.J ....

0.01

f - Frequency - Hz

~
I

-

II

Voo=3.3V
RL=10kn
Av=-1 VN
CB=1ILF

+

c
0

'E

~

.2
c

ill
Av=-6VN
0.1

0

i
:c
!

{!.

V

0.01

I

Z

+

AV=-2VN -

Q

:c

I-

II 1111

0.001
20

100

1k

10k 20k

0.001

L.....I....L..Ju..J..IW---L....J....J....L..u..u.L-...I-...L....I...I..L.........- " '

20

f - Frequency - Hz

1k

100

10k 20k

f - Frequency - Hz

Figure 7

FigureS

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-69

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S211C - AUGUST1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

10~~_

VOO=3.3V
RL= 10kn
AV=-1 VN
Ca=1j.LF

If!.
I

I

VOO=5V
RL=10kQ
Po = 300 j.LW -+-+-I-ttt+t---l-+-+-I-+t1Ht----t
Ca=1j.LF

+

I..

0.1

co

20Hz _

j

10 kHz

=I

r"'"
~

S
~

0.01

I

~

i!:

20Hz
1 kHz
I

0.001

I I

10

5

100

200

Po - Output Power - j.LW

f - Frequency - Hz

Figure 10

Figure 9
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10

10

If!.

If!.

VOO=5V
RL=10kn
Av=-1 VN
Ca=1j.LF

I

~

Z

+

+
c

C

0

0

t!

~
~0

t!
0

~

Po = 300 j.LW

0.1

i
:c
S

PO=200j.LW ~

"0.01

I

.'\

""~

~

D..

+

ii i

Q

0.001
20

~
'20

0.1
20Hz

i
:c
S
~

20kHz

'\

k--"

0.01

I

Z

i!:

VOO=5V
RL=10kn
AV=-1 VN
Ca = 1j.LF

I

3l
15
z

Z

+

10kHz

Q

:c

1
11111j.LW

I-

10k 20k

1k

100

,II

0.001
5

10

f - Frequency - Hz

100
Po - Output Power -j.LW

Figure 12

Figure 11

~TEXAS

INSTRUMENTS
3-70

1k' -

POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

t-

I I-t500

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211 C - AUGUST1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

'#

2

I!l

!§

FREQUENCY

I

PO=75mW
t= RL=SO
r-CB=111F

10

'#

AV=-o

./
./

j

s
~

z
0

'E

AV=-1

L

~

L

0.01

~

r- Po=15mW

..

N-

'"V

0

E

a

J:

S

~

0.01

~
I

Z

Z

Po=75mW

+

~

C

J:

J:

I-

0.001

100

20

1k

10k 20k

II III

0.001
100

20

f - Frequency - Hz

vs

OUTPUT POWER

FREQUENCY

'0
z
+
c

..
z
..
I

•

2~kHZ

....I

'E

1,9 kHz

0

'E

.~

W,I W~L 21~
v~~,*N

l2
~ ~~

,2

0.1

0

~

~v=

c

0

E

a

0.1

J:

~I

~

20 Hz
1--0

0.01

s

N

!::::V '"

0

1 kHz

S

Z
+
C
J:

E

0

~

i

r-

:::I

ii:
c

,~

VOO=5V
Po=100mW
RL=SO
CB=111F

=

'0

r

I 11111111

==
==

CD

I-

0

2

'#
Voo = 3.3 V
RL=sO
Av=-1 VN

OJ

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10
I
CD

10k 20k

Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

'#

1k
f - Frequency - Hz

Figure 13

J:

~

.A ~

0.1

'2

I

I-

Po=30mW

+

c

/~

0

'" '"

c

'0

~

c

.2

..
CD

:A;,j';'-2VN

0.1 r-..

Voo = 3.3 V
RL=SO
AV=-1 VN

I

1lllll

r-

ii:

ij

vs

FREQUENCY

~O~~I~.~I~

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

I
Z

.Ll

~

J:

I-

I-

0.01
10m

0.1

0.001

0.3

20

100

1k

10k 20k

f - Frequency - Hz

Po - Output Power - W

Figure 15

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-71

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211C - AUGUST1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

.,.

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

POWER OUTPUT

10

10
~VOO=5V
r:RL=SO
r- AV=-1 VN

I

Iz

VOO=5V
RL=SO
AV=-1 VN
Po=30mW

~

i

.So!

c

~~

0.1 :: PO=60mW

0

;---

E
:I!
!

~I

r-PO=10mW
0.001
100

1k

20Hz

,[1

~

II 1111
20

0.1
Po - Output Power - W

Figure 17

Figure 18

SUPPLY RIPPLE REJECTION RATIO

SUPPLY RIPPLE REJECTION RATIO

vs

vs

FREQUENCY

FREQUENCY

0

I

I
c

-10

1-1'0

r--....

-30

t

1:;:1'0
-40

a:

-so

'ii'

!D.
ii:

~
D.
D.

-60 ·1 B=

I...... It

I"

~

JI
1~1I=1 ~F
~

I' ~I-J

JJJs~lll ~ .65

-70

-90

"a

I
0

~

~

I/'

~~

-20
-30

·f
a:
!
I:

-50

:s

t"'--r-C;B= .1

~c

~
D.
D.

V

r-

VOO=5V
RL=SOto10kO -

-10
ID

~

r--..

I"

-80
~

I I

~B, = 1.1 ~F

r-...

:s

III

II

~
-40
-60 ·1 B=

~ i"""

l'~

~

I/'
I-!:::

-70

~ ~~

-80

III

1k

10k 20k

-100
20

f - Frequency - Hz

ii ITilil i
25
•

1k

100

f - Frequency - Hz

Figure 20

Figure 19

~TEXAS

INSTRUMENTS
3-72

I

.!

t'-....

t-...

I'" ~N

-90
100

~F

1~11=1 ~F

l"-

P

-100
20

...

0

VOO = 3.3 V
RL=SOto10kn -

-20

~

0.01
10m

10k 20k

f - Frequency - Hz

ID

} kHz

0.01

Z

"a

-

10kHz

I I

0.1

c+

i!=

I k~kHZ

-

I

+

c

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

10k 20k

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211 C - AUGUST1998 - REVISED MARCH 2000

TYpiCAL CHARACTERiSTiCS
OUTPUT NOISE VOLTAGE
vs
FREQUENCY

OUTPUT NOISE VOLTAGE
vs
FREQUENCY

20

~
I

20

10

>:::I.

GI
CI

10

I
GI
CI

~

.!

~

~

GI

z~

GI
UI

"0

z

'5
D'5

'5

.e-:::I

0

I

0

I III

.;
1
20

100

11['

I

VDD=3.3V
BW = 10 Hz to 22 kHz
AV=-1 VN
RL=8 Qto 10 kQ

>c

1k

1
20

10k 20k

VDD=5V
BW = 10 Hz to 22 kHz
RL = 8 Q to 10 kQ
~V=-11 VN
I

100

f - Frequency - Hz

1k

Figure 22

CROSSTALK
vs
FREQUENCY

CROSSTALK
vs
FREQUENCY

-so

-00
Po=25mW
VDD = 3.3 V
RL=32Q
CB=1I1F
AV=-1 VN

-70
-75
III

'D

...
!
I

UI

e
0

-80

-S5
-60
-65

i'

III

V

N2 00

-85

'D

..."iii
I

'Iii

-90
-95

UI

,~

-100

Lj...o
IN

-105
-110
20

V

]..IV
~

II

111 ~f

1k
f - Frequency - Hz

e
0

Ip~I~~I~m~
-

VDD=3.3V
RL=8Q
- CB=1I1 F
_, AV=-1 VN

-70
-75

~

-80

-85

I
100

10k 20k

f - Frequency - Hz

Figure 21

-65

I

~'-

II
II

10k 20k

IN2TOOUT

~

10-

1V
[..-

po
IN1TOOUT2

-90
-95
-100
20

100

1k

10k 20k

f - Frequency - Hz

Figure 23

Figure 24

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-73

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SL0S211C - AUGUSTl998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CROSSTALK

-60

vs

FREQUENCY

FREQUENCY

m
I

-60

()

-60

CB=1 J.1F
I- RL=8('!

AV=-1 VN
-65

I'

~

If

VOO=5V""

-55 I- PO= 100mW

-

-65

'Q

-50

VOO=5V "'
Po=25mW CB =1 J.1F
RL=32('!
AV=-1 VN

-65

-75

CROSSTALK

vs

m

I'

-85

IN2TO~UT1

-90

"- ~

-95

~

-100

l>

'r

'Q

r--

1

I---

-70

I

f

()

~~

III

-110
20

IN2TOOUT1
1\

f'

1.;'1-'

Vj-

-90
'NI1iIIIM -95

IIII!

100
1k
f - Frequency - Hz

-60
-85

IN1TOOUT2
-105

1'"

-75

-100
20

10k 20k

Figure 25

II ilill
100
1k
f - Frequency - Hz

10k 20k

Figure 26

MUTE ATTENUATION

MUTE ATTENUATION

vs

vs

FREQUENCY

FREQUENCY

0
-10
-20

m
'Q

I

VOO = 3.3 V
RL=32('!
CB=1 J.1F

f8

-30

-30 1--+-H-f-H++I--++++ltItt--t--H-tHHit---l

I

c

-40

ii
il!

g

-40~+-~H#~~+++Hm-~~~mr~

-60

11
il!

-60~+-~H##-~+++Hm-~~~mr~

i

-70

0

~
!I:E

~ -60~~~m=~+#~~~~~

-60
-70

-60

~+-++l+H+I--+-+++++H+--+-H+t-Htt--1

-601--+-~~~+-rH+ffl~+-rH~t-1

-90

-901--~HH+m~~YK+m~~YK+m~

-100
20

-100 L-..l-J....LL..U.LJ..L.....--LJ...l.J..UW--.l.....J...J..l..J..Il.I.LL:-....J
20
100
1k
10k 20k
f - Frequency - Hz

100

1k
f - Frequency - Hz

10k 20k

Figure 27

Figure 28

~TEXAS

3-74

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211C - AUGUSTl998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE MARGIN

vs
FREQUENCY
100

V~~1~13~ ~

~~

80

~

III

"cI

TA=25°C
No Load

'iii
CJ

Phase

~

Q.

40

0

120°

I I I" I

~

60

oS
~
!.
0

150°
I II

Gain

.5

90°

!:

~

,

:2

J

60°

I

20

E

30°

.e-

~
0

-20

0°

10

100

1k
10k
f - Frequency - Hz

-300
10M

100k

Figure 29
OPEN-LOOP GAIN AND PHASE MARGIN

vs
FREQUENCY
100

~~~~I~VI

,~
80
III

"c

I\~

=

TA 25°C
No load

'0;
Q.

1!.
0

Phase

~

CJ
0

I I I II

,~

60

I

40

Gain

,

120°

90°,5

i

~
60°

30°

r-....

0

1k

10k

100k

!i

~
I

20

-20
100

150°
II

1M

!

0°

-300
10M

f - Frequency - Hz

Figure 30

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

3-75

TPA122
1S0-mW STEREO AUDIO POWER AMPLIFIER
SLOS211C

AUGUST1998

REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

vs

LOAD RESISTANCE

LOAD RESISTANCE

120

100

~
I

J
:;
Q.
:;

0
I

,p

300

THO~N= 1 ~

l'1

VOO=3.3V
AV=-1VN

~

80

OUTPUT POWER

vs

-

250

7200

;o

"~

40

............

20

'"

D.

150

I

100

i,p

16

"- I'-..

56

o

64

8

16

24

vs

SUPPLY VOLTAGE

FREQUENCY

'#.

64

F

I

VI=1 V
AV=-1 VN
t- RL = 10 IUl
r- CB=1 flF

f:

!z

1.2

!

i5:
c

c(

E

~

I

0.1

i

0.&

(.)

()

j

0.6

::J

III

~

0.4

-L

0.01

S
~
I

0.2
0

~

~
j!:
2.5

3

3.5

4

4.5

5

5.5

0.001
20

Voo - Supply Voltage - V

Figure 33

100

1k
f - Frequency - Hz

Figure 34

~TEXAS

3-76

56

Q

TOTAL HARMONIC DISTORTION PLUS NOISE

~

I
Q
Q

48

vs
1.4

Q.
Q.

40

Figure 32

SUPPLY CURRENT

~

32

r--.r--

RL - Load Resistance -

Figure 31

I

~

50

24
32
40
48
RL - Load Resistance - Q

-

1\

I--

0
8

VoO=5V
AV=-1VN

\\

;:

~

60

THO~N=1 ~

1\

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

10k 20k

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211C - AUGUST1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SIGNAL-TO-NOISE RATIO

104

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

VOLTAGE GAIN

FREQUENCY

I

VI=1 V
III

\

"I
0

I

i

~

100

 10). For this reason a low-leakage
tantalum or ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the
capacitor should face the amplifier input in most applications, as the dc level there is held at Vool2, which is
likely higher than the source dc level. Please note that it is important to confirm the capacitor polarity in the
application.

power supply decoupling, Cs
The TPA122 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure that the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 IlF, placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 IlF or greater placed near
the power amplifier is recommended.

~TEXAS .
3-82

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211C - AUGUST199B - REVISED MARCH 2000

APPLICATION INFORMATION

mid rail bypass capacitor, CB
The mid rail bypass capacitor, CB, serves several important functions. During start-up, CB determines the rate
at which the amplifier starts up. This helps to push the start-up pop noise into the subaudible range (so low it
can not be heard). The second function is to reduce noise produced by the power supply caused by coupling
into the output drive signal. This noise is from the midrail generation circuit internal to the amplifier. The capaCitor
is fed from a 160-kQ source inside the amplifier. To keep the start-up pop as low as possible, the relationship
shown in equation 6 should be maintained.

(C B

x

1
160

<_1_
(C,R,)

(6)

kn) -

As an example, consider a circuit where CB is 1 IlF, C, is 1 IlF, and R, is 20 kQ. Inserting these values into the
equation 9 results in: 6.25:0; 50 which satisfies the rule. Bypass capacitor, CB, values of 0.1 IlF to 1 IlF ceramic
or tantalum low-ESR capaCitors are recommended for the best THD and noise performance.

output coupling capacitor, Cc
In the typical single-supply single-ended (SE) configuration, an output coupling capacitor (Cc) is required to
block the dc bias at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling
capacitor, the output coupling capaCitor and impedance of the load form a high-pass filter governed by
equation 7.
fc =

1
2:rtR L C C

(7)

The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the
low-frequency corner higher. Large values of Cc are required to pass low frequencies into the load. Consider
the example where a Cc of 68 IlF is chosen and loads vary from 32 Q to 47 kQ. Table 1 summarizes the
frequency response characteristics of each configuration.

Table 1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode
RL

Cc

LOWEST FREQUENCY

320

68 1lF

73Hz

10,0000

681lF

0.23 Hz

47,0000

681lF

0.05 Hz

As Table 1 indicates, headphone response is adequate and drive into line level inputs (a home stereo for
example) is very good.
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. With the rules described earlier still valid, add the following
relationship:
(8)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-83

TPA122
150-mW STEREO AUDIO POWER AMPLIFIER
SLOS211C - AUGUST1998 - REVIseD MARCH 2000

APPLICATION INFORMATION

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this application. A real capacitor can be modeled simply as
a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the beneficial effects
of the capacitor in the circuit. The lower the equivalent value of this resistance, the more the real capacitor
behaves like an ideal capacitor.

s-y versus 3.3-Y operation
The TPA122 was designed for operation over a supply range of 2.7 V to 5.5 V. This data sheet provides full
specifications for 5-V and 3.3-V operation since these are considered to be the two most common standard
voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain
setting, or stability. Supply current is slightly reduced from 3.5 rnA (typical) to 2.5 rnA (typical). The most
important consideration is that of output power. Each amplifier in the TPA122 can produce a maximum voltage
swing ofVoo - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as opposed
when VO(PP) = 4 V while operating at 5 V. The reduced voltage swing subsequently reduces maximum output
power into the load before distortion begins to become significant.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAu.AS, TEXAS 75265

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
o PACKAGE

• 300-mW Stereo Output
• PC Power Supply Compatibility S-V and
3.3-V Specified Operation
•
•
•
•
•

(TOP VIEW)

IN1

V01

Shutdown Control
Internal Mid-Rail Generation
Thermal and Short-Circuit Protection
Surface-Mount Packaging
Functional Equivalent of the LM4880

SHUTDOWN
BYPASS

GND
Voo
Vo2

IN2

description
The TPA302 is a stereo audio power amplifier capable of delivering 2S0 mW of continuous average power into
an 8-0 load at less than 0.06% THO +N from a S·V power supply or up to 300 mW at 1% THO+ N. The TPA302
has high current outputs for driving small unpowered speakers at 8 0 or headphones at 32 O. For headphone
applications driving 32-0 loads, the TPA302 delivers 60 mW of continuous average power at less than 0.06%
THO + N. The amplifier features a shutdown function for power-sensitive applications as well as internal thermal
and short·circuit protection. The amplifier is available in an 8-pin sOle (0) package that reduces board space
and facilitates automated assembly.

typical application circuit

VOO 6

RF

J-

Audio

Input

~

.L

RI

IN1

3

BYPASS

4

IN2

I
CI·

Audio

~C

L

RI

.,.

I

I
2

1

.A.

I:

V01

I:

V02 5

-::-

1

:(
Cc

CBT

Input

~

fes

VOol2

8

VOO

SHUTDOWN

I
I

t
Bias
Control

Ii

I~C

I~
7

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

~TEXAS

Copyright © 2000. Texas Instruments Incorporeted

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-85

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
SLOS174B -JANUARY 1997 - REVISE MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

SMALL OUTLINEt
(D)

-40°C to 85°C

TPA3020

t The 0 packages are available taped and reeled. To order a taped
and reeled part, add the suffix R (e.g., TPA3020R)

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)*
Supply voltage, VDD ....................................................................... 6 V
Input voltage, VI .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. -0.3 V to Voo + 0.3 V
Continuous total power dissipation .................... Internally Limited (See Dissipation Rating Table)
Operating junction temperature range, TJ .......................................... -40°C to 150° C
Storage temperature range, Tstg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
t Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

TA $ 25°C
POWER RATING

DERATING FACTOR
ABOVE TA 25°C

TA =70°C
POWER RATING

TA =85°C
POWER RATING

o

731 mW

5.8mWrC

460mW

380mW

=

recommended operating conditions
MIN

MAX

Supply voltage, VOO

2.7

5.5

UNrr
V

Operating free-air temperature, TA

-40

85

°c

dc electrical characteristics at specified free-air temperature, Voo = 3.3 V (unless otherwise noted)
PARAMETER
100

Supply current

Via

Input offset voltage

PSRR

Power supply rejection ratio

IOO(SO)

Quiescent current in shutdown

TEST CONDITION

MIN

VOO =3.2Vto 3.4 V

TYP

MAX

2.25

5

mA

5

20

mV

55

dB

0.6

ac operating characteristics, VOO

UNIT

20

I1A

=3.3 V, TA =25°C, RL =8 n (unless otherwise noted)

PARAMETER

TEST CONDITION

MIN

THO < 0.08%

TYP

MAX

UNIT

100

THO < 1%

125

Po

Output power

Gain =-1,
f= 1 kHz

BaM

Maximum output power bandwidth

Gain = 10,

20

kHz

Bl

Unity gain bandwidth

Open loop

1.5

MHz

Channel separation

f= 1 kHz

75

dB

Supply ripple rejection ratio

1= 1 kHz

45

dB

Noise output voltage

Gain =-1

10

j,LVrms

Vn

THO < 0.08%,

RL=320

25

THO < 1%,

RL=320

35

l%THO

~TEXAS

INSTRUMENTS
3-86

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

mW

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
SLOS174B- JANUARY 1997 - REVISE MARCH 2000

dc electrical characteristics at specified free-air temperature, Voo = 5 V (unless otherwise noted)
PARAMETER

TEST CONDITION

100

Supply current

Voo

Output offset vo~age

See Note 1

PSRR

Power supply rejection ratio

VDD =4.9Vto5.1 V

iDD(SD)

Quiescent current in shutdown

MIN

TYP

MAX

4

10

UNIT
rnA

5

20

mV

65

dB

0.6

,..A

ac operating characteristics, Voo = 5 V, TA = 25°C, RL = 8 Q (unless otherwise noted)
PARAMETER

TEST CONDITION

MIN

THO < 0.06%
Po

Output power

Gain =-1,
1= 1 kHz

BOM

Maximum output power bandwidth

Gain = 10,

Bl

Unity gain bandwidth

Open loop

Vn

TYP

MAX

UNIT

250

THO < 1%

300

THO <0.06%,

RL=32n

60

THO < 1%,

RL=32n

80

I%THD

mW

20

kHz

1.5

MHz

Channel separation

1= 1 kHz

75

dB

Supply ripple rejection ratio

1=1 kHz

45

dB

Noise output voltage

Gain =-1

10

IlVrms

typical application
RF

VDD
-:::-

Stereo Audio
Input

R~
-:::-

R,

8 IN13 BYPASS

C,

L~ ~,

Jl "-

RL

From Shutdown
Control Circuit (TPA4860)
R,

4 IN2-

5

Cc

RL

=
=

250 mW per Channel at RL 8 n
60 mW per Channel at RL 32 Q

RF

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-87

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
SLOS174B - JANUARY 1997 - REVISE MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE

THO+N

100
Vn

vs Frequency

1-3,7-9,
13-15,19-21

vs Output power

4-6,10-12
16-18,22-24

Total harmonic distortion plus noise

Supply current

vs Supply voltage
vs Free-air temperature

Output noise voltage

vs Frequency

Maximum pa~ge power dissipation

vs Free-air temperature

Power dissipation

vs Output power

POmax

Maximum output power

vs Free-air temperature

Po

Output power

vs Load resistance
vs Supply voltage

25
26
27,28
29
30,31
32,33
34

35
36
37
38,39
40,41

Open loop response
Closed loop response
Crosstalk

vs Frequency

Supply ripple rejection ratio

vs Frequency

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

'#.
I

I!!I
it

vs

vs

FREQUENCY

FREQUENCY

10
f=VCC=5V
~ Po = 250 mW
I-RL=80
I- AV=-5VN

!=VCC=5V
~ Po=250mW
I- RL=80
I- A =-1 VN

~

i

i

,

/

~

V02~

0.1 ~

i

{!.

V01:=

I

I.

~

l:::=

j
Z

+

+

Q

Q

0.010
20

100
1k
f - Frequency - Hz

10k 20k

i!:

0.010
20

100

1k

f - Frequency - Hz

Figure 2

Figure 1

~TEXAS

INSTRUMENTS
3-88

~V01

~

I

Z

i!:

./

V02

~

POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

10k 20 k

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
SLOS174B - JANUARY 1997 - REVISE MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

#.

OUTPUT POWER

10

I

#.

.1

t: Po = 250 mW

~

!I

it
c
o

I

V01

"U'

.2
c

~

:!

I'

1

i

is

I

u

V02 I--

I

'2
o

i

0.1

::t:

!

0.1

!

{!!.

{!!.

I

+

Q

IJ

V02

I

Z

+

~

~ V01

I

Z

j:

VCC=5V
f=20Hz
RL=8n
AV=-1 VN

3:
~
!I

-RL=8n
,-A =-10VN

it

10

I

I- VCC=5V

-I

Q

0.010
20

100
1k
f - Frequency - Hz

10 k 20 k

j:

0.010
0.01

0.1
Po - Output Power - W

Figure 3

Figure 4

TOTAL HARMONIC DISTORTION PLUS NOISE

#.

vs

OUTPUT POWER

OUTPUT POWER

10

#.
I

f= VCC=5V

I

.~
!I

it
c
o

10
~VCC=5V
I-f=20kHz
- RL=8n
I- AV=-1VN

3:

!=f=1kHz
I-RL=8Q
I-AV=-1VN

z

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

(5

z

!I

it

5

i5

~

i

V01

.2
c

u

l""I'+oI.

'2

'E
i5

i
!

0.1

~

E
t::::::

V01

-

V02

:::I.

>:::I.

I

I

CD

CD

CI

.!

CI

!

100

~

~

5:

.~
z0

'0

z

'5

'5

~

0

100

~

V01

10

0

10

I

I

c

V(J2

c

>

>

1
20

1k

100

10 k 20 k

f - Frequency - Hz

f - Frequency - Hz

Figure 28

Figure 27
MAXIMUM PACKAGE POWER DISSIPATION
vs
FREE-AIR TEMPERATURE

POWER DISSIPATION
vs
OUTPUT POWER
0.75
VOO=5V

iii:
I

c
0

ia.

0.75

Q

J

0.5

I

c

.ll!
()

0.5

ia.
'\

CD
CI

~
E
:::J
E

iii:

1'\I\.

'iii
.!!

~,

0.25

I

~

",
"

..

'R

0.25

-25

0

25
50
75
100 125 150
TA - Free-Air Temperature - °C

175

I

~=160

"

:IE

o

r

'iii

5

o

o

Figure 29

Two Channels Active

I
0.25
0.5
Po - Output Power - W

0.75

Figure 30

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-95

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
SLOS1748

JANUARY 1997 - REVISE MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION

MAXIMUM OUTPUT POWER

vs

vs

OUTPUT POWER

FREE·AIR TEMPERATURE

0.3

160

Voo = 3.3 V
Two Channels Active

140

0.25

;::
I

c

i

·iii
II)

120

0.2

0.15

L

0.1

I

11.

V-

0.05

o

t,

RL=16Q

1\ \

100

is

I

VOO=5V
Two Channels Active

---

tiQ

80

.'

60

~RL=16Q

~. RL=SQ

'--

40

o

0.05

0.1
0.15
0.2
0.25
Po - Output Power - W

0.3

20

0.35

o

0.25
0.5
Po max - Maximum Output Power -

Figure 31

OUTPUT POWER

vs

vs

FREE·AIR TEMPERATURE

LOAD RESISTANCE

150

P
I

e:::I

"Iii
~

a.

400

1\

J

350

RL=16Q

~/

130

E
~

i

120

LI..
I

;::
I

'"

I

200

~

150

0

Il.

250

11.

:;

\
J'\.."oo = 5 V

\

I

rP

110

100

100

0.075
0.15
Po max - Maximum Output Power - W

0.225

o

5

10

Figure 33

15
20 25 30
35 40
RL - Load Resistance - Q

Figure 34

~TEXAS

INSTRUMENTS
3-96

'"

r-....

"- ~=3l3V r-. i - '""""- ~

50

Voo = 3.3 V
Two Channels Active

o

1

300

E

RL=SQ

10). For this reason a low-leakage
tantalum or ceramic capaCitor is the best choice. When polarized capaCitors are used, the positive side of the
capacitor should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely
higher than the source dc level. Please note that it is important to confirm the capaCitor polarity in the application.

~TEXAS

3-100

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA302
300·mW STEREO AUDIO POWER AMPLIFIER
SLOS174B - JANUARY 1997 - REVISE MARCH 2000

APPLICATION INFORMATION
power supply decoupling, Cs
The TPA302 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure that the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 I1F, placed as close as possible to the device VDD lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 I1F or greater placed near
the power amplifier is recommended.

midrail bypass capacitor, CB
The midrail bypass capacitor, CB, serves several importantfunctions. During startup or recovery from shutdown
mode, CB determines the rate at which the amplifier starts up. This helps to push the start-up pop noise into
the subaudible range (so low it can not be heard). The second function is to reduce noise produced by the power
supply caused by coupling into the output drive signal. This noise is from the mid rail generation circuit internal
to the amplifier. The capacitor is fed from a 25-kn source inside the amplifier. To keep the start-up pop as low
as possible, the relationship shown in equation 6 should be maintained.

1
(C B x 25

<_1_

kn) -

(6)

(C,R,)

As an example, consider a circuit where CB is 0.1 I1F, C, is 0.22 I1F and R, is 10 kQ. Inserting these values into
the equation 9 results in: 400:0; 454 which satisfies the rule. Bypass capacitor, CB, values of 0.1 I1F to 1 I1F
ceramic or tantalum low-ESR capacitors are recommended for the best THO and noise performance.

output coupling capacitor, Cc
In the typical single-supply single-ended (SE) configuration, an output coupling capacitor (Cd is required to
block the dc bias at the output of the amplifier thus preventing dc currents in the load. As with the input coupling
capacitor, the output coupling capacitor and impedance of the load form a high-pass filter governed by
equation 7.
fc =

1
2:n:RL C c

(7)

The main disadvantage, from a performance standpoint, is that the load impedances are typically small, which
drives the low-frequency corner higher. Large values of Cc are required to pass low frequencies into the load.
Consider the example where a Cc of 68 I1F is chosen and loads vary from 8 n, 32 n, and 47 kn. Table 1
summarizes the frequency response characteristics of each configuration.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-101

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
SLOS1?48 - JANUARY 1997 - REVISE MARCH 2000

APPLICATION INFORMATION

Table 1. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode
Cc

LOWEST FREQUENCY

80

6811F

293 Hz

320

6811F

73Hz

47,0000

6811F

0.05 Hz

RL

As Table 1 indicates, most of the bass response is attenuated into 8-0 loads while headphone response is
adequate and drive into line level inputs (a home stereo for example) is very good.
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the following relationship:

(c s x

1
25

<_1_~_1_

kn) - (CIR I)

RLeC

(8)

shutdown mode
The TPA302 employs a shutdown mode of operation designed to reduce quiescent supply current, IOO(q)' to
the absolute minimum level during periods of nonuse for battery-power conservation. For example, during
device sleep modes or when other audio-drive currents are used (Le., headphone mode), the speaker drive is
not required. The SHUTDOWN input terminal should be held low during normal operation when the amplifier
is in use. Pulling SHUTDOWN high causes the outputs to mute and the amplifier to enter a low-current state,
100 < 1 !lA. SHUTDOWN should never be left unconnected because amplifier operation would be unpredictable.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real capacitor can be modeled
simply as a resistor in series with an ideal capacitor. The voltage drop across tliis resistor minimizes the
beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the
real capacitor behaves like an ideal capacitor.

~TEXAS

INSTRUMENTS
3-102

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

TPA302
300-mW STEREO AUDIO POWER AMPLIFIER
SLOS174B-JANUARY 1997 - REVISE MARCH 2000

APPLICATiON INfoRMATiON
thermal considerations
A prime consideration when designing an audio amplifier circuit is internal power dissipation in the device. The
curve in Figure 43 provides an easy way to determine what output power can be expecfed out of the TPA302
for a given system ambient temperature in designs using 5-V supplies. This curve assumes no forced airflow
or additional heat sinking.
160

VDD=5V
TlNo Channels Active

140
I

if

120

t,

RL= 160

" '-\

100 ~,RL=80 f"

(!!.

80
60

40
20

o

0.25

0.5

0.75

Po max - Maximum Output Power - W

Figure 43. Free-Air Temperature Versus Maximum Output Power

5-V versus 3.3-V operation
The TPA302 was designed for operation over a supply range of 2.7 V to 5.5 V. This data sheet provides full
speCifications for 5-V and 3.3-V operation since are considered to be the two most common standard voltages.
There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or
stability. Supply current is slightly reduced from 3.5 rnA (typical) to 2.5 mA (typical). The most important
consideration is that of output power. Each amplifier in the TPA302 can produce a maximum voltage swing of
Voo - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) 2.3 V as opposed when
VO(PP) = 4 V while operating at 5 V. The reduced voltage swing subsequently reduces maximum output power
into the load before distortion begins to become significant.

=

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-103

3-104

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
o OR OGN PACKAGE
(TOP VIEW)

• Fully Specified for 3.3-V and 5-V Operation
• Wide Power Supply Compatibility
2.SV-5.5V

=
=

VO-

SHUTDOWN
BYPASS

=

• Output Power for RL 8 n
- 350 mW at Voo 5 V, BTL
- 250 mW at Voo 3.3 V, BTL
• Ultra-Low Quiescent Current in Shutdown
Mode ••• 0.15IlA
• Thermal and Short-Circuit Protection
• Surface-Mount Packaging
- SOIC
- PowerPADTM MSOP

GND

Voo
Vo+

description
The TPA301 is a bridge-tied load (BTL) audio power amplifier developed especially for low-voltage applications
where internal speakers are required. Operating with a 3.3-V supply, the TPA301 can deliver 250-mW of
continuous power into a BTL 8-n load at less than 1% THD+N throughout voice band frequencies. Although
this device is characterized out to 20 kHz, its operation was optimized for narrower band applications such as
cellular communications. The BTL configuration eliminates the need for external coupling capacitors on the
output in most applications, which is particularly important for small battery· powered eqUipment. This device
features a shutdown mode for power-sensitive applications with a quiescent current of 0.15!lA during shutdown.
The TPA301 is available in an 8-pin sOle surface-mount package and the surface-mount PowerPAD MSOP,
which reduces board space by 50% and height by 40%.

VOO
RF

l

Audio
Input

~C

R,

~
I

CB
O.1I1F

4

IN-

3

IN+

2

BYPASS

r

,
,
,
,
,
,
,
,
,
,
,
,

-=:::-

-=

.A.
.a.

1

.k Cs
T 111F
-=

Voot2

T

From System Control

6

SHUTOOWN

I
I

---~~

V

'VV"

L-~
r

-

~

: Y

Bias
Control

VOO

VO+ 5

J

1

I

Vrr

8~
.........

350mW

7
GNO

---:b

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments IncOrporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3-105

TPA301.
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARYl998 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES

MSOP
Symbolization

TA

SMALL OUTLINEt
(D)

MSOpt
(DGN)

-40°C to 85°C

TPA3010

TPA3010GN

AAA

t The 0 and OGN packages are available taped and reeled. To order a taped and reeled part, add
the suffix R to the part number (e.g., TPA301 DR).

Terminal Functions
TERMINAL
NAME

NO.

1/0

DESCRIPTION

I

BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected
to a O.l-I1F to l-I1F capacitor when used as an audio amplifier.

BYPASS

2

GNO

7

IN-

4

I

IN+

3

I

IN + is the noninverting input. IN + is typically tied to the BYPASS terminal.

SHUTDOWN

1

I

SHUTDOWN places the entire device in shutdown mode when held high (100 < 1 ItA).

GNO is the ground connection.
IN- is the inverting input. IN- is typically used as the audio input terminal.

VOO

6

VO+

5

0

VO+ is the positive BTL output.

Vcr

8

0

Vcr is the negative BTL output.

VOO is the supply voltage terminal.

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)*
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
:(: Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE

=

PACKAGE

TAS25°C

DERATING FACTOR

0

725mW

5.8mWfOC

TA 70°C
464mW

OGN

2.14W§

17.1 mWfOC

1.37W

TA

=85°C

377mW
1.11 W

§ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
Operating free-air temperature, TA

:ilTEXAS
3-106

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

MIN

MAX

2.5

5.5

UNIT
V

-40

85

°C

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

=

=

electrical characteristics at specified free-air temperature, Voo 3.3 V, TA 25 u C (uniess otherwise
noted)
PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

5

20

mV

0.7

1.5

rnA

0.15

5

j.lA

VOO

Differential output voltage

See Note 1

PSRR

Power supply rejection ratio

VOO = 3.2 V to 3.4 V

85

100Cal

Supply current (see Figure 3)

BTL mode

100(sd)

Supply current, shutdown mode (see Figure 4)

UNIT
dB

NOTE 1: At 3 V < VOO < 5 V the dc output voltage is approximately Vool2.

operating characteristics, Voo

= 3.3 V, TA = 25°C, RL = 8 n

PARAMETER
Po
THO+N

B,

Vn

TEST CONDITIONS

Output power, see Note 2

THO = 0.5%,

See Figure 9

Total harmonic distortion plus noise

Po =250 mW,
Gain=2,

f= 20 Hz to 4 kHz,
See Figure 7

Maximum output power bandwidth

Gain =2,
See Figure 7

THO = 3%,

Unity·gain bandwidth

Open Loop,

See Figure 15

Supply ripple rejection ratio

f= 1 kHz,
See Figure 2

CB=Ij.lF,

Noise output voltage

Gain = 1,
RL=32Q,

CB=O.Ij.lF,
See Figure 19

MIN

TYP

MAX

250

UNIT
mW

1.3%
10

kHz

1.4

MHz

71

dB

15

j.lV(rms)

NOTE 2: Output power is measured at the output terminals 01 the device at 1= 1 kHz.

electrical characteristics at specified free-air temperature, Voo = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER
VOO

Oifferential output voltage

PSRR

Power supply rejection ratio

100(a)

Quiescent current (see Figure 3)

TEST CONDITIONS

MIN

THO+N

B,

Vn

20

UNIT
mV

0.7

1.5

rnA

0.15

5

j.lA

dB

= 5 V, TA = 25°C, RL = 8 n

PARAMETER
Po

MAX

5
78

VOO=4.9Vt05.1 V

10Q(sd} Quiescent current, shutdown mode (see Figure 4)

operating characteristics, Voo

TVP

TEST CONDITIONS

MIN

TVP

Output power

THO =0.5%,

See Figure 13

700

Total harmonic distortion plus noise

Po =250 mW,
Gain =2,

1= 20 Hz to 4 kHz,
See Figure 11

1%

Maximum output power bandwidth

Gain =2,
See Figure 11

THO=2%,

Unity-gain bandwidth

Open Loop,

See Figure 16

Supply ripple rejection ratio

f= 1 kHz,
See Figure 2

CB= lj.lF,

Noise output voltage

Gain = 1.
RL=32n,

CB = O.Ij.lF,
See Figure 20

MAX

UNIT
mW

10

kHz

1.4

MHz

65

dB

15

j.lV(rms)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-107

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

VDD 6

.r

RF

~c

RI

I
I

I

4

IN-

3

IN+

2

BYPASS

,r
,
,
,
,
,
,
,
,
,
,
,
,
,
,

CB -:::::0.1 IlF

T
-=-

1

::L

VDoJ2

-

Audio
Input

SHUTDOWN

J-

I
I

---.~
V

T-=-

VO+ 5

,A

RL=8

,A

~~
r

-

Vo- 8

•

:V
I

Bias
Control

7

GND

n-

Figure 1. Test Circuit

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
Supply voltage rejection ratio

vs Frequency

IDD

Supply current

vs Supply voltage

3,4

Output power

'vs Supply voltage

5

Po

vs Load resistance

Total harmonic distortion plus noise

vs Frequency
vs Output power

6
7,8,11,12
9,10,13,14

Open loop gain and phase

vs Frequency

15,16

Closed loop gain and phase

VS

Frequency

17,18

Vn

Output noise voltage

vs Frequency

1~,20

PD

Power dissipation

vs Output power

21,22

~TEXAS

INSTRUMENTS
3-108

2

kSVR

THD+N

VDD
Cs
11lF

POST OFFICE BOX 655:303 • DALLAS, TEXAS 75265

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS
SUPPLY VOLTAGE REJECTION RATIO

SUPPLY CURRENT

vs
FREQUENCY

SUPPLY VOLTAGE

vs

o

!8

-10

I

.2

~
c

j

1.1
RL=8Q
CB= 11lF
0.9

-20
c(

E

-30

-

I

-40

C
~
:::I

0.7

~
r::L
r::L
:::I

0.5

(J

1"-60 I--

VOO=5V

--

11

-.llWll

bo""

-70

"

-I-"

III
I

"iT
is"

Voo = 3.3 V

0.3

E

-80

0.1

-90
-100
20

100

~.1

10 k 20 k

1k

2

4

3

f - Frequency - Hz

5

6

VOO - Supply Voltage - V

Figure 2

Figure 3
SUPPLY CURRENT (SHUTDOWN)

vs
SUPPLY VOLTAGE
0.5
SHUTOOWN

=High

0.45
c(
:::l.
I

0.4

C

0.35

I!!

~

(J

>-

8:
:::I

0.3
0.25

III

Is
~

E

0.2

,.,

0.15
0.1
0.05
2

2.5

3

3.5

4

4.5

5

5.5

VOO - Supply Voltage - V

Figure 4

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-109

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
SUPPLY VOLTAGE
1000
THD+N1%
800

~I

I

RL/f

'5

~

0

/

/

600

1/

400

J

/
./

I

~
200

V"

RL= 32

. . .V

'YV"

............. 1'

f..--'
o

2

2.5

3
3.5
4
4.5
VDD - Supply Voltage - V

5

5.5

Figure 5
OUTPUT POWER
vs
LOAD RESISTANCE
800
700

~

800

i

500

i

400

1\

~
r\.VDD=5 V

I

i.
0

I

~

THD+N=1%

300

'\.
r-....

200

100

o

"

~DD=3.3V

8

16

......

~

I'....

............

r--

-- -

24
32
40
48
RL - Load Resistance - 0

56

Figure 6

~TEXAS

INSTRUMENTS
3-110

POST OFFICE BOX 655303 • DAu.AS. TEXAS 75265

64

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10

~

vs

FREQUENCY

FREQUENCY

1=

VOO=3.3V
~ Po=250mW : AV=-~~
t- RL=8n
III

I

.~
Z

~

c
0

'f

~

~

J:

0.1

~

I"-

""'~

~
AV =-2 VN

~

111

III
Po =,50 mW

~

"'"

L

~ RL=8n
_ AV=-2VN

+
c

~

AV=-.10V~ ~

0

~ VOO=3.3V

~
Z

/

0

:!i
Q

10

I

v::V

+

.2
c

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

~

~0

f--

~
J:

'!

iii

I
Z
+
Q

'T

..L.

-

Po=125mW -

~~

0.1

0

~

J:

I-

0.01

I'

Q

"'"

i=
100

20

~

z+

1k

10k

20k

Po = 250 mW
0.01
20

f - Frequency - Hz

rr-

31

'0

z

vs

OUTPUT POWER

OUTPUT POWER
~

Voo=3.3V
f=1 kHz
AV=-2VN

..

/

f=20kHz

"0

/

+

z

;:

r--- I-

+

c

I

0

0

~

'Iii

f=1kHz

..
c

is

.2
c

RL' 8n

0

0

~

0.1

'!

f= 10kH;-

I" I

0

;:

0

~
J:

10

I

c

20k

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

f:::

10k

FigureS

TOTAL HARMONIC DISTORTION PLUS NOISE

I

1k
f - Frequency - Hz

Figure 7

~

100

-

/

0.1

iii

~

;2I

+

Z

I
Z

f=20Hz -;;;;;

+
Q

Q

J:

i=

I-

0.01
0.04

0.1

0.16

0.22

0.28

0.34

0.4

0.01
0.01

Po - Output Power - W

VOO=3.3V
RL=8n
AV=-2VN

I lLllll
0.1
Po - Output Power - W

Figure 9

Figure 10

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-111

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10

'#.

vs

FREQUENCY

FREQUENCY
10

1=

1=

::
'0
+

c
0

'E
0

S
.~

AV =-10 VN ./

0

i

0.1

~

r-...t--

VOO=5V
~ RL=8Q
r- AV=-2VN

::
i+

./

0

i

~

.~
0

..

~
:c

AV=-2VN

!

Z
+
C

c+

~I

PO=175mW ~,

0.1

~
I

"'I

Z

:c

i!:

I-

0.01

100

20

1k

10k

20k

I~

prii~lill

0.01
20

100

f - Frequency - Hz

10

vs

OUTPUT POWER

OUTPUT POWER
10

r-

'#.

..

VOO=5V
I-- f= 1 kHz
I-- AV=-2VN

~

Z

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

f=

I

-

I

.~

/

+

~

~

+

~0

0

II

ic

f=1kHz

'E

I

r--

.g

c
0

.~

E

0.1

!

!

0.1 ~ f=20Hz

~

VOO~5V

!

~I

I

Z
+
C

Z
+
C

:c

RL=8Q
AV =-2 VN

:c

I-

0.01
0.1

I-

0.25

0.40

0.55

0.70

0.85

0.01
0.01

Po - Output Power - W

I I I
0.1
Po - Output Power - W

Figure 14

Figure 13

~TEXAS

3--112

I---J
f= 10 kHz

c

RL=8Q
./

is

f=20kHz

10...

z

/

c

20k

Figure 12

TOTAL HARMONIC DISTORTION PLUS NOISE

'#.

10k

1k
f - Frequency - Hz

Figure 11

..

II':

~~Io"

'E

!

:c

V
po=~mw

c

/
~

~

F

I

ra.i-'
/ V'

z

:c

'#.

VOO=5V
PO=350mW : AV=-20VN
t- RL=8Q
. '-

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

r-

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY

40

RL=Open

30
Gain
III
'1:1

20

c

'iii
0

120

~~

~r--..

~

I

180

V~O=3,3V

t-K~~~~~

60

10

I

a.
0
0

....

..c

o
0

D..

r\

a.

0

-10

-60

-120

-20
-30

j

104

1

-180

f - Frequency - kHz

Figure 15
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY

40

tbf~~~

30
Gain
III

'1:1
I

c

'iii
0

20

180
VOO=5V
RL=Open

~,
\.

10

120

'

60

......

0
0

....

..

c

...

0

I

a.

0

\.

0

D..

\

a.
0 -10

.c

-60

-120

-20
-30

104

1

-180

f - Frequency - kHz

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-113

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS20BC - JANUARY199B - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

V

0.75
0.5
!XI

"cI

iii

CJ

I

0.25

!~

.........

180
~

/

\

-0.25

(

-0.5

/

170

\

I

0

Q.

g

Phase

160

"'\

Gain

-0.75
-1
VDD=3.3V
RL=80
Po = 0.25 W
CI=lI1F

-1.25
-1.5

-2
101

I

150

1\

102

104

1lI

l

140

1\
\

--I

-1.75

o

\
\
\

130

106

120

f - Frequency - Hz

Figure 17
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

V

0.75
0.5
!XI

"I
c

~Q.
0

i
~

0.25

I

0

I

......

\

170

\

Gain

I

-0.5

180

r--....

/

/'

-0.25

Phase

,\,

160
o

\

-0.75

140

-1
-1.25
-1.5
-1.75

1\

VDD=5V
RL=80
PO=O.35W
CI=lI1F

1\
\,

-----r

130
120

-2
101

106
f - Frequency - Hz

Figure 18

~1ExAs

3-114

I
I

150

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARYl998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

100

OUTPUT NOISE VOLTAGE

vs

vs

FREQUENCY

FREQUENCY
100

Voo=:UV
BW = 22 Hz to 22 kHz
RL=32Q
CB=O.l",F
AV=-l VN

Ui'

~

::!.

I

~

::!.

I

CD
CI

VOBTL

ll!
!t

~

.~

10

z
:;
a.
:;

VO+

:;
a.
:;

II

CD
CI

VOBTL

ll!

~z

VOO=5V
BW = 22 Hz to 22 kHz
RL=32Q
CB=O.l",F
AV=-l VN

Ui'

0

I illil

11

10

Vo+

0

I

I

c

>c

>

1
20

100

1k

10 k

1
20

20k

100

f - Frequency - Hz

Figure 19

vs

OUTPUT POWER

OUTPUT POWER

/

I

240

L
L

c

ia.

210

·iii
.!!
Q

I

D.
I

,p

180
150
120

---...

..-

/

720
640

==E
I

o

/

560

i

J

480

/

"iii

is

I

I
:.

I

,p

I

VOO=3.3V
RL=8Q
-

100

200

300
Po - Output Power - mW

400

.....

V

V

c

-I

90

20k

POWER DISSIPATION

vs
300

==E

10k

Figure 20

POWER DISSIPATION

270

1k
f - Frequency - Hz

400

7

320

Voo=5V
RL=8Q

240
160

o

_

I
200

400

600

800

1000

1200

Po - Output Power - mW

Figure 21

Figure 22

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-115

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARYl998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridge-tied load
Figure 23 shows a linear audio power amplifier (APA) in a BTL configuration. TheTPA301 BTL amplifier consists
of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive
configuration but power to the load should be initially considered. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing do.."n, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load: Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 1).
V

_ VO(PP)
(rms) -

2/2

2
V(rms)

(1 )

Power - - - -

RL

Voo

J'

RL

~

J'!
rv ~

VO(PP)

2x vO(PP)

-VO(PP)

Figure 23. Bridge-Tied Load Configuration
In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an

8-0 speaker from a single-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is
a 6-dB improvement - which is loudness that can be heard. In addition to increased power, there are frequency
response concerns. Consider the single-supply SE configuration shown in Figure 24. A coupling capaCitor is
required to block the dc offset voltage from reaching the load. These capacitors can be quite large
(approximately 33 /J.F to 1000 /J.F) so they tend to be expensive, heavy, occupy valuable PCB area, and have
the additional drawback of limiting low-frequency performance of the system. This frequency limiting effect is
due to the high pass filter network created with the speaker impedance and the coupling capacitance and is
calculated with equation 2.

~TEXAS

3-116

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SL0S208C - JANUARY1998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridge-tied load versus single-ended mode (continued)
f

(2)

1
(comer) - 2:n:R LC c

For example, a 68-~F capacitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the de offsets, eliminating the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

Voo

J'

~dB~----~~====

;VO(PP)

C~R}J'; v"'PP)
fe

Figure 24. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased intemal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of a SE
configuration. Intemal dissipation versus output power is discussed further in the thermal considerations
section.

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these ineffiCiencies is voltage drop across the
output stage transistors. There are two components of the intemal voltage drop. One is the headroom or de
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from Voo.
The intemal voltage drop multiplied by the RMS value of the supply current, loorms, determines the intemal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in
the amplifier, the current and voltage waveform shapes must first be understood (see Figure 25).
'00

,/

V(LRMS)

-~-

'OO(RMS)

Figure 25. Voltage and Current Waveforms for BTL Amplifiers

~TEXAS

INSTRUMENTS
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3-117

TPA301
3SD-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

P
Efficiency = ~

(3)

SUP

Where:

PL
VLrms

= vLrms 2 =
RL

=

Vp

12

Psup
loorms

V 2
-p2RL

V DD loorms

=

Voo 2Vp
Jt RL

2Vp
RL

Jt

Jt

Efficiency of a BTL Configuration =

Jt

p R
( --'=-..b

)1/2

2

Vp

W-

(4)

oo
Table 1 employs equation 4 to calculate efficiencies for three different output power levels. The efficiency of the
amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in a
nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output
power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper
power supply design.
Table 1. Efficiency vs Output Power in 3.3-V 8-Q BTL Systems

(W)

EFFICIENCY
(%)

PEAK-lo-PEAK
VOLTAGE
(V)

INTERNAL
DISSIPATION

0.125

33.6

1.41

0.26

OUTPUT POWER

47.6
2.00
0.25
2.45t
58.3
0.375
t High-peak voltage values cause the THO to increase.

(W)

0.29
0.28

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 4, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

~TEXAS

3-118

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

APPLICATION INFORMATION

application schematic
Figure 26 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of
-10 VN.

VDD 6

,-~AA-----'---YVv--------------~---+--

VDD/2

Audio
Input

©----J
':(
~

CI
RI
0.4711F 10 lin

4

IN-

3

IN+

2

BYPASS

__-----VDD
Cs

T

111F

VO+ 5

Vo- 8

350mW

7
GND
From System Control

1

SHUTDOWN

Figure 26. TPA301 Application Circuit
The following sections discuss the selection of the components used in Figure 26.

component selection
gain setting resistors, RF and RI
The gain for each audio input of the TPA301 is set by resistors AF and AI according to equation 5 for BTL mode.
BTL Gain

= Av = -

2(~~)

(5)

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA301 is a MOS amplifier, the input impedance is very high,
consequently input leakage currents are not generally a concern although noise in the circuit increases as the
value of AF increases. In addition, a certain range of AF values are required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kQ and 20 kQ. The effective impedance is calculated in equation 6.
Effective Impedance

=

AA
~

A F
F+

(6)

I

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-119

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SL0S208C - JANUARY1998 - REVISED MARCH 2000

APPLICATION INFORMATION
component selection (continued)
As an example, consider an input resistance of 10 kil and a feedback resistor of 50 kil. The BTL gain of the
amplifier would be -1 0 VN, and the effective impedance at the inverting terminal would be 8.3 kil, which is well
within the recommended range.
For high performance applications metal film resistors are recommended because they tend to have lower noise
levels than carbon resistors. For values of RF above 50 kil the amplifier tends to become unstable due to a pole
formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor, CF, of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kn. This, in effect, creates a low-pass filter network with the cutoff frequency defined in equation 7.

~dBF=====~~-----(7)

fCo(lowpasS)

fCO

For example, if RF is 100 kn and CF is 5 pF then fco is 318 kHz, which is well outside of the audio range.

input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.

fco(highpass) =

23t~ICI

(8)

The value of C, is important to consider as it directly affects the bass (lOW frequency) performance of the circuit.
Consider the example where RI is 10 kil and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
(9)

~TEXAS

3-120

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

APPLICATION INFORMATION

component selection (continued)
In this example, C, is 0.40 I1F so one would likely choose a value in the range of 0.47 I1F to 1 11F. A further
consideration for this capacitor is the leakage path from the input source through the input network (R" C,) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications, as the dc level there is held at Vool2, which is likely higher
than the source dc level. It is important to confirm the capacitor polarity in the application.
power supply decoupling, Cs
The TPA301 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also prevents
oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved
by using two capacitors of different types that target different types of noise on the power supply leads. For
higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 I1F, placed as close as possible to the device Voo lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 I1F or greater placed near the audio
power amplifier is recommended.
midrail bypass capacitor, CB
The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. Ouring
start-up or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and THO
+ N. The capacitor is fed from a 250-kn source inside the amplifier. To keep the start-up pop as low as pOSSible,
the relationship .shown in equation 10 should be maintained, which insures the input capaCitor is fully charged
before the bypass capacitor is fully charged and the amplifier starts up.
10
(C B x 250

<

1

(10)

kn) - (RF + RI) CI

As an example, consider a circuit where CB is 2.2I1F, C, is 0.47I1F, RF is 50 kn and R, is 10 kn. Inserting these
values into the equation 10 we get:
18.2

~

35.5

which satisfies the rule. Bypass capacitor, CB, values of 2.211F to 1 I1F ceramic or tantalum low-ESR capacitors
are recommended for the best THO and noise performance.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this application. A real (as opposed to ideal) capacitor can
be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes
the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance, the more
the real capacitor behaves like an ideal capacitor.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-121

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SLOS208C - JANUARY1998 - REVISED MARCH 2000

APPLICATION INFORMATION

s-y versus 3.3-Y operation
The TPA301 operates over a supply range of 2.5 V to 5.5 V. This da~a sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability.
The most important consideration is that of output power. Each amplifier in TPA301 can produce a maximum
voltage swing of VOO - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) 2.3 V as
opposed to VO(PP) = 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into
an 8-n load before distortion becomes significant.

=

Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power for a given output-power level than operation from 5-V supplies.

headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA301 data sheet, one can see that when the TPA301
is operating from a 5-V supply into a 8-n speaker 350 mW peaks are available. Converting watts to dB:
P dB = 10LogPW = 10Log 3500 mW = -4.6 dB
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
-4.6 dB - 15 dB

- 19.6 dB (15 dB headroom)

-4.6 dB - 12 dB

- 16.6 dB (12 dB headroom)

-4.6 dB - 9 dB

- 13.6 dB (9 dB headroom)

-4.6 dB - 6 dB

- 10.6 dB (6 dB headroom)

-4.6 dB - 3 dB

- 7.6 dB (3 dB headroom)

. Converting dB back into watts:
Pw

1QPdB/10
11 mW (15 dB headroom)
= 22 mW (12 dB headroom)
= 44 mW (9 dB headroom)
= 88 mW (6 dB headroom)
= 175 mW (3 dB headroom)

~1EXAS

3-122

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA301
350-mW MONO AUDIO POWER AMPLIFIER
SL0S208C - JANUARY1998 - REVISED MARCH 2000

APPLICATION INFORMATION

headroom and thermal consideratIons (continued)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 350 mW of continuous power output with 0 dB
of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings
for the system. Using the power dissipation curves for a 5-V, 8-0 system, the internal dissipation in the TPA301
and maximum ambient temperatures is shown in Table 2.
Table 2. TPA301 Power Rating, 5-V, 8-0, BTL
MAXIMUM AMBIENT
TEMPERATURE

PEAK OUTPUT POWER
(mW)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(mW)

350

350mW

600

46°C

350

175 mW (3 dB)

500

64°C

350

88 mW (6 dB)

380

85°C

350

44 mW (9 dB)

300

98°C

350

22mW(12dB)

200

115°C

350

11 mW (15 dB)

180

119°C

OCFM

Table 2 shows that the TPA301 can be used to its full 350-mW rating without any heat sinking in still air up to
46°C.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-123

3-124

TPA311
350·mW MONO AUDIO POWER AMPLIFIER
• Fully Specified for 3.3·V and 5-V Operation
• Wide Power Supply Compatibility
2.5V-5.5 V

=

• Output Power for RL 8 a
- 350 mW at Voo = 5 V, BTL
- 250 mW at Voo = 5 V, SE
- 250 mW at Voo 3.3 V, BTL
- 75 mWat Voo 3.3 V, SE

=
=

• Shutdown Control
- 100 7 itA at 3.3 V
- 100 = 60 !1A at 5 V
• BTL to SE Mode Control
• Integrated Depop Circuitry
• Thermal and Short-Circuit Protection
• Surface Mount Packaging
- SOIC
- PowerPADTM MSOP

=

D OR DGN PACKAGE
(TOP VIEW)

description

The TPA311 is a bridge-tied load (BTL) or
SHUTDOWN
Vosingle-ended (SE) audio power amplifier develBYPASS
GND
oped especially for low-voltage applications
SElBTL
VDD
where internal speakers and external earphone
IN
Vo+
operation are required. Operating with a 3.3-V
supply, the TPA311 can deliver 250-mW of
continuous power into a BTL 8-0 load at less than 1% THD+N throughout voice band frequencies. Although
this device is characterized out to 20 kHz, its operation was optimized for narrower band applications such as
cellular communications. The BTL configuration eliminates the need for external coupling capacitors on the
output in most applications, which is particularly important for small battery-powered eqUipment. A unique
feature of the TPA311 is that it allows the amplifier to switch from BTL to SE on the fly when an earphone drive
is required. This eliminates complicated mechanical switching or auxiliary devices just to drive the external load.
This device features a shutdown mode for power-sensitive applications with special de pop circuitry to virtually
eliminate speaker noise when exiting shutdown mode and during power cycling. The TPA311 is available in an
8-pin sOle surface-mount package and the surface-mount PowerPAD MSOP, which reduces board space by
50% and height by 40%.

VDD 6
RF

~CI

VDoJ2

-=-

Audio
Input
RI

-=-

4

IN

2

BYPASS

CBFT

-=SHUTDOWN

From System Control
From HPJack

3

Vcr 8

SE/BTL

7
GND

.A.
..m.

Please be aware that an important notice concerning availability. standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS .
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

Copyright © 2000, Texas InstrumentS Incorporated

3-125

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B.,. JANUARY 1998 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES

MSOP
Symbolization

TA

SMALL OUTLINEt
(D)

MSOpt
(DGN)

-40°C to 85°C

TPA3110

TPA3110GN

AAB

t The 0 and OGN packages are available taped and reeled. To order a taped and reeled part, add
the suffix R to the part number (e.g., TPA311 DR).

Terminal Functions
TERMINAL
NAME

NO.

BYPASS

I/O

DESCRIPTION

I

BYPASS is the tap to the voltage divider for intemal mid·supply bias. This terminal should be connected
to a O.l-I1F to 1-I1F capacitor when used as an audio amplifier.

2

GNO

7

IN

4

GNO is the ground connection.
I

IN is the audio input terminal.

3

I

When SElBTL is held low, the TPA311 is in BTL mode. When SElBTL is held high, the TPA311 is in SE
mode.

SHUTDOWN

1

I

SHUTDOWN places the entire device in shutdown mode when held high (100 =60 /lA, VOO =.5 V).

VOO

6

VO+

5

yO"'"

8

SElBTL

VOO is the supply voltage terminal.

0
0

VO+ is the posHive output for BTL and SE modes.
V0"'" is the negative output in BTL mode and a high-impedance output in SE mode.

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)*
Supply voltage; Voo ....................................................................... 6 V
Input voltage, VI ........................................... __ ................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

:f: Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other condHions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

TAS;25°C

DERATING FACTOR

TA = 70°C

TA = 85°C

0

725mW

5.8mWI"C

464mW

3nmW

OGN

2.14W§

17.1 mWI"C

1.37W

1.11 W

§ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
Operating free-air temperature, TA (see Table 3)

~TEXAS

3-126

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

MIN

MAX

2.5

5.5

V

-40

85

°C

UNIT

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SL0S207B - JANUARY 1998 - REVISED MARCH 2000

=

=

electrical characteristics at specified free-air temperature, Voo 3.3 V, TA 25°C (unless otherwise
noted)
PARAMETER

VOO

TEST CONDITIONS

Output offset voltage (measured differentially)

PSRR

Power supply rejection ratio

100

Supply current (see Figure 6)

IOO(SO)

Supply current, shutdown mode (see Figure 7)

MIN

See Note 1
VOO = 3.2 V to 3.4 V

IBTL mode
ISE mode

TYP

MAX

5

20

85

UNIT

mV
dB

83

BTL mode

0.7

1.5

SEmode

0.35

0.75

7

50

rnA

J.LA

NOTE 1: At 3 V < VOO < 5 V the dc output voltage is approximately VOO/2.

operating characteristics, Voo

=3.3 V, TA =25°C, RL =8 n

PARAMETER

TEST CONDITIONS

THO =0.5%,

BTL mode,

THO =0.5%,

SEmode

MIN

See Figure 14

Po

Output power, see Note 2

THO+N

Total harmonic distortion plus
noise

Po=250mW,
See Figure 12

1 = 20 Hz to 4 kHz,

Gain =2,

BOM

Maximum output power bandwidth

Gain =2,

THO =3%,

See Figure 12

Bl

Unity-gain bandwidth

Open Loop,

See Figure 36

1= 1 kHz,
See Figure 5

CB=II1F,

BTL mode,

1= 1 kHz,
See Figure 3

CB=II1F,

SEmode,

Gain = 1,
BTL,

CB = O.II1F,
See Figure 42

RL=320,

Supply ripple rejection ratio

Vn

Noise output voltage

TYP

250
110

MAX

UNIT

mW

1.3%
10

kHz

1.4

MHz

71
dB
86
15

I1V(rms)

NOTE 2: Output power is measured at the output terminals 01 the device at 1 = 1 kHz.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-127

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, VDD
noted)

TEST CONomONS

PARAMETER
VOO

=5 V, TA =25°C (unless otherwise

PSRR

Power supply rejection ratio

100

Supply current (see Figure 6)

IOO(SO)

Supply current, shutdown mode (see Figure 7)

operating characteristics, VDD

VOO=4.9Vt05.1 V

I BTL mode

UNIT
mV
dB

76

BTL mode

0.7

1.5

0.35

0.75

60

100

TEST CONDITIONS
THO = 0.5%,

BTL mode,

THO = 0.5%,

SEmode

MIN
See Figure 18

THO+N

Total harmonic distortion plus
noise

Po = 350 mW,
See Figure 16

1= 20 Hz to 4 kHz,

Gain=2,

BOM

Maximum output power bandwidth

Gain =2,

THO = 2%,

See Figure 16

B1

Unity-gain bandwidth

Open Loop,

See Figure 37

1= 1 kHz,
See Figure 5

CB=1ILF,

BTL mode,

1=1 kHz,
See Figure 4

CB=1ILF,

SEmode,

Gain=1,
BTL,

CB=0.1ILF,
See Figure 43

RL = 32 Q,

TYP
700
300

NOTE 2: Output power is measured at the output terminals 01 the device at 1 = 1 kHz.

~TEXAS

INSTRUMENTS
3-128

20

SEmode

Output power, see Note 2

Noise output voltage

MAX

5
78

I SEmode

Po

Supply ripple rejection ratio

TYP

mA

IIA

=5 V, TA =25°C, RL =8 Q

PARAMETER

Vn

MIN

Output offset voltage (measured differentially)

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

MAX

UNIT
mW

1%
10

kHz

1.4

MHz

65
dB
75
15

ILV(rms)

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SL0S207B - JANUARY 1998 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

VDD 6

Ir-~~----'-~~--------------r--+--~~---VDD

-:!:-

RF

Audio
Input

C6H~1 ,~I

~

4

IN

2

BYPASS

~ Cs
T
1 "F
...

VDD/2

c

___ •

~>--e_____V_O_++-5~~---,-::-

r--~~~~--~".~V
-::k.

CB
0.1 I1F

T

L.-~

:
:

1

SHUTDOWN
_
3' SElBTL

I

-:!::-

I

r

-

V(T" 8

>--e-------t--------'

...

::-V
I

Bias
Control

l
Figure 1. BTL Mode Test Circuit

VDD 6
RF
Audio
Input

~C

RI

L

4

J

.

~

IN

I

c
, --, ~V
,
,
,
,
,
,
,
,
,
, c -L.-~
•
,
,
,

I
2
CB
0.1 I1F

BYPASS

-::k.

T
-::-

Jo-

VDD

~

VDD/2

1

SHUTDOWN

3

SElBTL

:-V

I Control
Bias

I

T-::VO+ 5

VDD
Cs
111F

'q
I'

Cc
33OI1F

RL =32n

-::-

V(T" B

7
GND

~

Figure 2. SE Mode Test Circuit

~TEXAS

INSTRUMENTS
POST OFACE BOX 655303 • DAUAS, TEXAS 75265

3-129

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE

100

vs Frequency

Supply current

vs Supply voltage

6,7

vs Supply voltage

8,9

Output power

Po

THO+N

3,4,5

Supply voltage rejection ratio

vs Load resistance

10,11

vs Frequency

12,13,16,17,20,
21, 24, 25, 28, 29,
32,33

vs Output power

14,15, 18, 19,22,
23,26,27, 30, 31,
34,35

Total harmonic distortion plus noise

Open loop gain and phase

vs Frequency

36,37

Closed loop gain and phase

vs Frequency

38,39,40,41
42,43

Vn

Output noise voltage

vs Frequency

Po

Power dissipation

vs Output power

44, 45, 46, 47

TYPICAL CHARACTERISTICS

0

m

"0
I

I
c

t
a:
J
'ii'

~
~
a.
a.
::s

III

,

SUPPLY VOLTAGE REJECTION RATIO
vs
FREQUENCY

SE

-20
-30

I\..

-50
-60

~

-70
-80

Blii~~ = 1j VIII

r- I
20

100

",
..........: ~

ta:

-30

-50

III
til

-60
-70

::s

-60

~

I-

"-

~

.........:::; ~

-+WJII

-100
20

./

I I """ I III
II IIII
100

f - Frequency - Hz

Figure 3

~TEXAS

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

"

II
1k
f - Frequency - Hz

Figure 4

INSTRUMENTS
3-130

~

B~~i~t' 1/2 V

III

10 k 20k

j.lF

r-...

-90 r-

1k

~CB=0.1

CB = 11J.F.....

~

~
a.

~

-40

'ii'
~

/
J

VOO=5V
RL=SO
SE

-20

0

ic
/

CB=1!!F ....

-100

m

r-...

,

-10
"0
I

I\.. CB = 0.11J.F

~

-40

-90

0

Voo = 3.3 V
RL=SO

-10

SUPPLY VOLTAGE REJECTION RATIO
vs
FREQUENCY

10 k 20 k

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY CURRENT
vs
SUPPLY VOLTAGE

SUPPLY VOLTAGE REJECTION RATIO
vs
FREQUENCY

1.1

0
RL=8Q
CB =111F
BTL

-10

m
'a

-20

I

J
c

~GI

'Gi'

ex:

GI

'"

~

E
I

C
~
:s

-40

-50
~

-70

I:L
I:L

-80

:s

'

t- BTL

+
c

'#.

+
c

III
Po=50mW

-

i'-~

0

~

-

'E
0

~
~0

..

E

:c

0.1

"

;y

t'

~

~

AV=-10~

AV =-2 VN

0

E
01

"'"

~

~

.~

r--

:c

Po=125mW -

~~

0.1

li
~

I

I

Z

+

I'.

C

...

:c

r~

Z
+
C

...

:c

0.01
20

100

1k

10k

20k

Po=250 mW
0.01
20

Figure 12

Figure 13

vs

OUTPUT POWER

OUTPUT POWER
10

'#.

=

VOO=3.3V
f=1 kHz
f- AV=-2VN
t- BTL

~
+
c

Iz

./

/

r---- t-

+
c

j

0

'E

.s

I--

f= 10 kHz-

/

I III

0

~

III

is
u

C

f=1 kHz

r--

.5:!

c

RL' 80

0

.E

f= 20 kHz

I

~

:c

20k

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

1=

Z

10k

1k
f - Frequency - Hz

10
I

100

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

'#.

'"

0

5

0.1

:c

li

li

~

0.1

~I

I

Z
+
C

f=20Hz -

Z
+
C

...

:c

j:

0.01
0.04

0.1

0.16

0.22

0.28

0.34

0.4

0.01
0.01

Po - Output Power - W

VOO=3.3V
RL=80
AV =-2 VN
BTL

0.1
Po - Output Power - W

Figure 14

Figure 15

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-133

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

il-

~ VOO=5V
~ Po=350mW : AV=-20VN
I- RL=8Q
."
_ BTL
r.,

I

.~

Z

+

/

C

~

il-

.!:!
c

....

/

0

j

0.1

iii

t}- "'"

'I'--

~ RL=8Q
I- AV=-2VN
r- BTL

Z

+

~

~
.~

:J:

!

I
Z

i!:

C
:J:
I-

~

0.1

+

0.01

100

20

1k

10k

pril~ilil

0.01

20k

100

20

, - Frequency - Hz

10
I

vs

OUTPUT POWER

OUTPUT POWER
10

=
VOO=5V
f=1 kHz
-

il-

+

f=20kHz

I

AV=-2VN
BTL

.;

/

/

C

~
~

RL=8Q
./

is
.!:!
c

~

+

i---

c

,= 10 kHz

~0

~

"

'=1 kHz

I

r--

.~

0

r

0

Ii!01

0.1

:J:

!

!

~

0.1 t;::: f=20Hz

~

I

I

Z

Z
+
C

+

C
:J:
I-

0.01
0.1

:J:
I-

0.25

0.40

0.55

20k

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

~

~
Z

10k

Figure 17

TOTAL HARMONIC DISTORTION PLUS NOISE

il-

1k
f - Frequency - Hz

Figure 16

0.70

0.85

0.01
0.01

VOo=5V
RL=8Q
AV =-2 VN
BTL

Po - Output Power - W

0.1
Po - Output Power - W

Figure 18

Figure 19

~TEXAS

3-134

Pd=175mW -

~

Z

:J:

--

V __

0

Ii

AV =-2 VN

I"""

~~

0

'f!
0

;2I

Ii

V

po=~mw

C

1/
~

AV =-10 VN

~ VOO=5V

.;0

~

is

10

I

INSTRUMENTS
POST OFACE BOX 655303 • DALlAS. TEXAS 75265

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10

#.
I

vs

FREQUENCY

FREQUENCY
10

F YOO=3.3Y

z

+
c

Gl
III

.~

Z

+
c

"""V

0.1

0.01

~

..
!..
:c

I~P"

0.1

0

~

"-

Po ~10mW ..4

0

'2

Ay=-1 YN

!01

Ay=-10YN -

l.......- i'

I

j

I I

I

Ay=-5YN

+

C

IIIII

I-

0.001
20

1k

100

I-

10k

0.001

20k

PO' :15mW

r-;1t,~~,iW

:c

I

~

1=

Z

+
C

:c

~

;~

0.01

{l.

I

Z

20

vs

OUTPUT POWER

OUTPUT POWER
10

1=

:

'0
+

c
0

=

YOO=3.3Y
RL=32Q
Ay=-1 YN ---,

I
Gl

/

~

-

+
c

i-

Z

l/'

;:

;:

~

..

/

~0

j

f= 10kHz

0

~

f=20kHz

SE

0

0

,

=

#.

YOO=3.3Y
t-- f=1 kHz
t-- RL=32Q
t-- Ay=-1 YN
SE

20k

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

z

10k

Figure 21

TOTAL HARMONIC DISTORTION PLUS NOISE

I

1k
f - Frequency - Hz

Figure 20

#.

I III

100

f - Frequency - Hz

Ii
:c

~

0

6'

:c
{l.

RL=32Q
Ay=-1 YN
SE

;:

0

j

r--

'0

~

i

1=

I

_ RL=32Q
SE

'0

F YOO=3.3Y

#.

~ PO=30mW

Gl
III

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

'2
0

/

0.1

!01

:c

j

j

{l.

{l.

Z
+
C

Z
+
C

I

I

:c

:c

I-

0.01
0.02

0.1

E
r--~

-

I-

0.025

0.03

0.035

0.04

0.045

0.05

J

f=1 kHz

0.01
0.002

f=20Hz

1 I

Po - Output Power - W

0.01

0.02 0.03

0.05

Po - Output Power - W

Figure 22

Figure 23

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-135

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10

'IJ!.

vs

FREQUENCY

FREQUENCY
10

1=

CD

.!!!

z0

+

I

V

AV=-10VN

~

",

0.1 ~ AV=-5VN

PO=15mW

i!
is

/

.S!
c

~01

{].

SE

+.

~~

0.1

0

I§

:z:

i

t- AV=-1 VN

c

~
~

~ RL=320

~
Z

LL

c

i

F VOO=5V

'IJ!.

VDO=5V
~ Po=60mW
t- RL=320
SE

I

TOTAL HARMONIC DISTORTION PLUS NOISE

VB

~l

:I!

~

/'"

0.01

AV=-1 VN

I

pO~'t30mW
jIP'

WM~

0.01

I

z

Z

~

Po=60mW

~

j!:

j!:

0.001
20

100

1k

10k

II

0.001

20k

100

20

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10

10

~

Voo=5V
t- 1=1 kHz
t- RL=320
I- AV=-1 VN
SE

I
+

c
0

'IJ!.

,

I

~Z

1/

i

~

I--

+

I

'E

~

~

c

II

0

i

Ii

{].

t-- 1=1 kHz
0.1

t:=

{].

I

I

Z

+

Q

...

:z:

j!:
0.04

0.06

0.08

0.1

0.12

0.14

0.01
0.002

Po - Output Power - W

VOO=5V
RL=320
AV=-1 VN
SE
0.01
Po - Output Power - W

Figure 27

Figure 26

~TEXAS

3-136

-

1=20 Hz

Z

+

Q

0.01
0.02

I

f=10kHz

.2

1/
0.1

n

Q

I

i

1=20 kHz

c

I

~0

i
:z:

20k

Figure 25

TOTAL HARMONIC DISTORTION PLUS NOISE

I

10k

f - Frequency - Hz

Figure 24

'IJ!.

""

1k

f - Frequency - Hz

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

0.1

0.2

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS2078 - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

.,.

.'"
I

vs

FREQUENCY

FREQUENCY

F VOO=3.3V

.,.

'- SE

Z

t- Po = 0.1 mW
t- RL=10kO

'0

z

~

A

~

.~

1

0.1

0

Ii!01

r-

AV=-1 VN

S

t-

.I~~=-~~NI

:z:

~
I

Z
+
0

t-

j!:

Wl1

100

0

u

1~~I=o.~mw

Ii

~

S

~

~

pr=IO.~ I~~II-

I
Z

+

0

111

1k

10k

0.01

20k

20

1k

100

vs

OUTPUT POWER

OUTPUT POWER

.,.

VOO=3.3V
f= 1 kHz
RL=10kO
AV=-1 VN

+
c

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

~

"6

I

~ SE

z

+

c

i:

cu

~

0
0

0.1

£

c

0

~

VOO=3.3V
RL=10kO
AV=-1 VN
SE

..'"

I

~

IS

10

I

~
C

20 k

Figure 29

TOTAL HARMONIC DISTORTION PLUS NOISE

Z

10k

f - Frequency - Hz

Figure 28

I

IIIII

:z:
I-

f - Frequency - Hz

.,.

/

1\

0.1

C

"i

1111

i:
0
1D
C

----,
I

:z:

II

Av=-5VN

20

".Po = 0.13mW

+
c

0

1.1

Uill

0.01

VOO=3.3V
RL=10kO
AV=-1VN
SE

I

+
c

i

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

0.1

I=l= f = 20 Hz

f=2OkHz

0

Ii!01

I

:z:

I

0.01

S

~

I

0.01

I
+

Z
+
0

1=1 kHz

Z
0

:z:
I-

:z:
I-

0.001
50

75

100

125

150

175

200

f= 10 kHz
I
I I I

0.001

5

Po - Output Power - I1W

10

=c:::::: ~

111
100

500

Po - Output Power -I1W

Figure 30

Figure 31

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-137

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

i=

fit.
I

J
+

VB

FREQUENCY

FREQUENCY

VDD=5V

fit.

r- PO=0.3mW
r- RL= 10 kn
r- SE

!z

VDD=5V
RL=10kn
AV=-1 VN

I

SE

+

c

I
.!!

/)

0.1

~

~

.!!

J

c0

IH

7z
~

I IIII I~

0.1

Po =0.2 mW

7'

~

~I;

AV=-2VN

Po=O.3mW -

E
:!

AV=-1 VN

S

i!:

TOTAL HARMONIC DISTORTION PLUS NOISE

VB

I

Po = 0.1 mW

Z

+

Q

:c

AV=-5VN

I-

0.01
20

100

1k

10k

20k

111111

0.01
20

100

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

VB

VB

OUTPUT POWER

OUTPUT POWER

10
VDD=5V
1=1 kHz
RL=10kn
AV=-1 VN

!z

+
c

fit.

~0

{].

VDD=5V
RL = 10 kn
Av=-1 VN

!z

I

I

~ SE

~
~

f = 20\kHZ _

is

<>

l

f=20Hz

0.1

E
as

I

I

If

I

0.01

z

Z

~
:c

~

i!:

I-

0.001
50

125

200

275

350

425

500

1=1 kHz
f= 10kHz
I I 11111

0.001

5

Po - output Power - I1W

10

100
Po - Output Power -I1W

Figure 34

Figure 35

~TEXAS

3-138

-

~ t-=

:c

/

0.01

SE

+
c

I

0.1

~
:c
S

10

I

~

~

20k

Figure 33

TOTAL HARMONIC DISTORTION PLUS NOISE

I

10k

f - Frequency - Hz

Figure 32

fit.

1k

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

-

I I
500

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN·LOOP GAIN AND PHASE

vs
FREQUENCY

40

~~~~

30
Gain'"
ID

'U

,

"

'\
10

a.

o

I

o

\.

0

8-

0

60

,

0

....0
c

120

....

\.

20

I

iCJ

180

VOO=3.3V
RL = Open
BTL

\

-10

-120

-20
-30

J

-60

102

1

104

-180

f - Frequency - kHz

Figure 36
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY
40

180

~~~~

VDo=5V
RL=Open
BTL

30

Gain
ID

'U
I

c
'ii
CJ

20

\:' ....
'\

10

c

0

r-

'\

0

0

III

:I
.c
D.

'\

8-

0

I- 60
I

a.
0

.9

' ....

I- 120

-10

l- -60

I- -120

-20
-30
1

103

101

104

-180

f - Frequancy - kHz

Figure 37

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-139

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

III

'OJ

"
0
0

j
0

-0.5

\

/'

,

-0.75
-1

-1.5
-1.75

VOO=3.3V
RL=SQ
Po = O.25W
CI=lILF
BTL

o

I

150

1\

102

\
loS

104

81

f.

140

1\
\

J

-2
101

160

\\
\
\
\

Gain

J

-1.25

170

\

I

0

Q.

""-

/

0.25

-0.25

Phase-""

/

0.5

"cI

180

V

0.75

130

106

120

f - Frequency - Hz

Figure 38
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

V"

0.75
0.5
III

"cI

OJ

"

0.25
0
-0.25

Q.

0

j..

G

-0.5

I

/

I

180

Phase ........ .........

/

\

/'

Gain

170

\
\

""\

o

\\

I

150

-0.75

140

-1
-1.25
-1.5
-1.75
-2
101

VOO=5V
RL=SQ
Po = 0.35W
CI=1ILF
BTL

103

\
\

130

loS

120
106

f - Frequency - Hz

Figure 39

~TEXAS

3-140

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

•

f.

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY
7
6

4

/1
1/

3

,

5
ID

'1::J

I

c
'OJ

"

a.

0
0

Phase

,

180

~

Gain

I

-

170

-

160

-

150
o

I
I

2

~

~

"

/'"

0

-1

-2

140
VOO=3,3V
RL=32Q
AV =-2 VN
Po=30mW
CI=l f,lF
Cc =470 f,lF
SE

-

130

-

120

-

110

~

-3

106

101

100

f - Frequency - Hz

Figure 40
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY
7
6
ID

5

'1::J

I

c

4

'iii

".3a.

3

If
1/

v-

Phase
Gain

180

~

I

-

160

-

150

-

140

-

130

-

120

-~

110

I

.c

2

=
0

"

170

0

0

-b

-

0

-1

VOO=5V
RL=32Q
AV=-2VN
Po =60 mW
CI=l f,lF
Cc =470 f,lF
SE

,

-2

106

101

=

II.

100

f - Frequency - Hz

Figure 41

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-141

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SL0S207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

OUTPUT NOISE VOLTAGE

va

va

FREQUENCY
100

'iii'

E

~
I

FREQUENCY
100

Voo=UV
BW = 22 Hz to 22 kHz
RL=32C
CB=O·l pF
t-AV=-l VN

E

;;;:::!.
I

t

OIl

II

~

l...u...w

CI

VOBTL

~

VOBTL

~

Iz

10

••z

VOO=5V
BW = 22 Hz to 22 kHz
RL=32C
CB=O.l pF
AV=-l VN

'iii'

VO+

i

11

10

Vo+

'S

f

0

0

.§'

.§'

I

I

1
20

100

1k

10k

1
20

20 k

100

f - Frequency - Hz

Figure 42

va

OUTPUT POWER

OUTPUT POWER

---

300

~
I

240

c

i

~

I
I

210

180
150

~

/

/

i--""

V

72

...

!it

E
I

Iis

1

I
I

VOO=3.3V
RL=8C
BTL

120

90

80

~

I

o

64

/

56

C

,

100

200
300
Po - Output Power - mW

400

48

40
32

/

~Jc

~

/

1

24

16

RL=32C
.~ ..... ........

V~1l =3.3V

8

o

SE

o

Figure 44

30
60
90
Po - Output Power - mW

Figure 45

~TEXAS

3-142

20k

POWER DISSIPATION

va

./

10k

Figure 43

POWER DISSIPATION

270

1k
f - Frequency - Hz

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

120

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION
vs
OUTPUT POWER

POWER DISSIPATION
vs
OUTPUT POWER

720

180

640

==E
I

/

560

c

i

/

480

I

0
II.
I

,p

400
320
240

160

160

V

140

i'ii

120

is

I

II.
I

80

VOO=5V
RL=80
BTL

200

400

600

800

1000

_

RL=80

II
Ir

K=320

II.

60
1200

V

I
/

100

Q

II
o

~I
c

/
/
I

OJ
.!!
Q

.....

V

40

o

VOO=5V
SE

I
50

Po - Output Power - mW

100

150

200

250

300

Po - Output Power - mW

Figure 46

Figure 47

APPLICATION INFORMATION

bridge-tied load versus single-ended mode
Figure 48 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA311 BTL amplifier consists
of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive
configuration but initially consider power to the load. The differential drive to the speaker means that as one side
is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the
load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where voltage is
squared, yields 4x the output power from the same supply rail and load impedance (see equation 1).
V

_ VO(PP)
(rms) 2/2

2
V(rms)

(1 )

Power - - - -

RL

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3--143

TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
bridge-tied load versus single-ended mode (continued)
voo

V' ;

RL

J'!
'V;

VO(PP)

2xVO(PP)

-VO(PP)

Figure 48. Bridge-Tied Load Configuration
In typical portable handheld equipment, a sound channel operating at 3.3 V and using bridging raises the power
into an 8-Q speaker from a single-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In terms of sound
power that is a 6-dB improvement, which is loudness that can be heard. In addition to increased power there
are frequency response concerns. Consider the single-supply SE configuration shown in Figure 49. A coupling
capacitor is required to block the dc offset voltage from reaching the load. These capacitors can be quite large
(approximately 33 !1F to 1000 !1F), tend to be expensive, heavy, and occupy valuable PCB area. These
capacitors also have the additional drawback of limiting low-frequency performance of the system. This
frequency limiting effect is due to the high-pass filter network created with the speaker impedance and the
coupling capacitance and is calculated with equation 2.

~=

00

1

23tR L C C

For example, a 68-IlF capacitor with an 8-Q speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

~TEXAS

3-144

INSTRUMENTS
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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridge-tied load versus single-ended mode (continued)

Voo

~dB~-----J~=====

Figure 49. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable, considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the thermal considerations
section.

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from Voo.
The internal voltage drop multiplied by the RMS value of the supply current, IOorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in
the amplifier, the current and voltage waveform shapes must first be understood (see Figure 50).
100

,/

V(LRMS)

-~-

IOO(RMS)

Figure 50. Voltage and Current Waveforms for BTL Amplifiers

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350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform, both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
PL

Efficiency = - -

(3)

Psup

Voo loorms
2Vp
RL

=

:It

:It

Efficiency of a BTL Configuration

:It

(

Vp

= W-

p R

--'=2-'=

)1/2
(4)

oo

Table 1 employs equation 4 to calculate efficiencies for three different output power levels. The efficiency of the
amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting in a
nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output
power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper
power supply design.
Table 1. Efficiency Vs Output Power in 3.3-V 8-Q BTL Systems
OUTPUT POWER

t

(W)

EFFICIENCY
(%)

0.125
0.25

33.6
47.6

PEAK-TO-PEAK
VOLTAGE

INTERNAL
DISSIPATION

(V)

(W)

1.41

0.26
0.29

2.00

2.45t
58.3
0.375
High-peak voltage values cause the THO to increase.

0.28

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. In equation 4, Voo is in the denominator. This indicates
that as Voo goes down,.efficiency goes up.

~TEXAS

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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

application schematic
Figure 51 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of

-10VN.

VDD 6

r-~~----~---V~--------------;----+----~~---VDD

VDoJ2

Audio

Input

RI

~

'T
-=-

10kQ

CI

0.47~F

Cc

4

IN

2

BYPASS

VO+ 5

330~F

T-=-

Cs

1~F

1 kQ

CB

2.2~FT

Vo- 8
7
1

From System Control

3
0.1 ~FT

SHUTDOWN r--"--..,
Bias
SElBTL
Control

GND

100kQ

VDD----~v-~------------------------------------------------~

100kQ

Figure 51. TPA311 Application Circuit
The following sections discuss the selection of the components used in Figure 51.

component selection
gain setting resistors, RF and RI
The gain for each audio input of the TPA311 is set by resistors RF and RI according to equation 5 for BTL mode.
BTL Gain

=

Av

= -

2(~~)

(5)

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA311 is a MOS amplifier, the input impedance is very high,
consequently input leakage currents are not generally a concern, although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kQ and 20 kQ. The effective impedance is calculated in equation 6.
Effective Impedance

RFRI
= =--'-:--:-RF

+ RI

-!!1
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(6)

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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SL0S2078 - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

component selection (continued)
As an example consider an input resistance of 10 kn and a feedback resistor of 50 k.Q. The BTL gain of the
amplifier would be -1 0 VN and the effective impedance at the inverting terminal would be 8.3 kn, which is well
within the recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kn the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor, CFo of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kn. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.

~dBF=====~~----fe(IOwpass)

(7)

fe

For example, if RF is 100 kn and CF is 5 pF then fe is 318 kHz, which is well outside of the audio range.
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.

fe(highpasS)

= 23t~ICI

(8)

The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kn and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
(9)

~TEXAS

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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

component selection (continued)
In this example, CI is 0.40 IlF, so one would likely choose a value in the range of 0.47 IlF to 1 IlF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI' CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage atthe inputto the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. It is important to confirm the capacitor polarity in the application.
power supply decoupling, Cs
The TPA311 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also prevents
oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved
by using two capacitors of different types that target different types of noise on the power supply leads. For
higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 IlF placed as close as possible to the device Voo lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 IlF or greater placed near the audio
power amplifier is recommended.
midrail bypass capacitor, CB
The midrail bypass capacitor, Ca, is the most critical capacitor and serves several important functions. Ouring
start-up or recovery from shutdown mode, Ca determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the mid rail generation circuit internal to the amplifier, which appears as degraded PSRR and
THO + N. The capacitor is fed from a 250-kn source inside the amplifier. To keep the start-up pop as low as
pOSSible, the relationship shown in equation 10 should be maintained, which insures the input capacitor is fully
charged before the bypass capacitor is fuly charged and the amplifier starts up.
10
(C a x 250

<

kn) -

1
(RF + RI) CI

(10)

As an example, consider a circuit where Ca is 2.2 IlF, CI is 0.47 IlF, RF is 50 kn and RI is 10 kn. Inserting these
values into the equation 10 we get: 18.2::; 35.5 which satisfies the rule. Bypass capacitor, Ca, values of 0.1 IlF
to 2.2 IlF ceramic or tantalum low-ESR capacitors are recommended for the best THO and noise performance.

single-ended operation
In SE mode (see Figure 51), the load is driven from the primary amplifier output (Vo+, terminal 5).
In SE mode the gain is set by the RF and RI resistors and is shown in equation 11. Since the inverting amplifier
is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included.
SE Gain

=

Av

= -

(~~)

(11 )

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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

single-ended operation (continued)
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the following relationship:
10
(C B x 250

~_1_

1

<

kn) -

(RF

+ R I)

CI

(12)

RLC C

As an example, consider a circuit where CB is 0.2.2IlF, CI is 0.47IlF, Cc is 330 IlF, RF is 50 knRL is 32 n, and
RI is 10 kn. Inserting these values into the equation 12 we get:
18.2 < 35.5 ~ 94.7 which satisfies the rule.

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 13.

(13)

fC(high pass)

The main disadvantage, from a performance standpoint, is that the typically small load impedances drive the
low-frequency corner higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 IlF is chosen and loads vary from 8 0,
32 0, to 47 kn. Table 2 summarizes the frequency response characteristics of each configuration.

Table 2. Common Load Impedances vs Low Frequency Output Characteristics in SE Mode
Cc

LOWEST FREQUENCY

SO

33OI1F

60Hz

320

330l1F

15Hz

47,0000

33Ol1F

0.01 Hz

RL

As Table 2 indicates an 8-0 load is adequate, earphone response is good, and drive into line level inputs (a home
stereo for example) is exceptional.

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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

SE/BTL operation
The ability of the TPA311 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional earphone amplifier in applications where
internal speakers are driven in BTL mode but external earphone or speaker must be accommodated. Internal
to the TPA311 , two separate amplifiers drive Vo+ and Vo-. The SElBTL input (terminal 3) controls the operation
ofthe follower amplifier that drives Vo- (terminal 8). When SElBTL is held low, the amplifier is on and the TPA311
is in the BTL mode. When SE/BTL is held high, the Vo- amplifier is in a high output impedance state, which
configures the TPA311 as an SE driver from VO+ (terminalS). 100 is reduced by approximately one-half in SE
mode. Control of the SElBTL input can be from a logic-level TTL source or, more typically, from a resistor divider
network as shown in Figure 52.

Cc
4

IN

2

BYPASS

Vo+ 5

330ILF

Vrr 8
1 SHUTDOWN
3
O.1ILF

T

SElBTL

.--.1............,

7

GND

Bias
Control

100kO

VDD----VV~~------------------------------------------------~

100kO

Figure 52. TPA311 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) mono earphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-kn/1-k.Q divider pulls the SElBTL input low. When a plug is inserted, the 1-k.Q
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the Vo- amplifier is
shutdown causing the BTL speaker to mute (virtually open-circuits the speaker). The Vo+ amplifier then drives
through the output capacitor (Cc) into the earphone jack.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this application. A real (as opposed to ideal) capacitor can
be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes
the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more
the real capacitor behaves like an ideal capaCitor.

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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
5-Y versus 3.3-Y operation
The TPA311 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability.
The most important consideration is that of output power. Each amplifier in TPA311 can produce a maximum
voltage swing of Voo - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) =2.3 V as
opposed to VO(PP) =4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into
an 8-0 load before distortion becomes significant.
Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power for a given output-power level of operation from 5-V supplies.

headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA311 data sheet, one can see that when the TPA311
is operating from a 5-V supply into a 8-0 speaker that 350 mW peaks are available. Converting watts to dB:

=

10Log (P w)
P ref

= 10Log

(35~ ;JW)

= -4.6 dB

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
-4.6 dB - 15 dB = - 19.6 dB (15 dB headroom)
-4.6 dB - 12 dB = - 16.6 dB (12 dB headroom)
-4.6 dB - 9 dB

- 13.6 dB (9 dB headroom)

-4.6 dB - 6 dB = - 10.6 dB (6 dB headroom)
-4.6 dB - 3 dB

= -

7.6 dB (3 dB headroom)

Converting dB back into watts:
Pw

=

10PdB/10 x Pref

= 11 mW (15 dB headroom)
= 22 mW (12 dB headroom)
= 44 mW (9 dB headroom)
= 88 mW (6 dB headroom)
= 175 mW (3 dB headroom)

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TPA311
350-mW MONO AUDIO POWER AMPLIFIER
SLOS207B - JANUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

headroom and thermal considerations (continued)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 350 mW of continuous power output with 0 dB
of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings
for the system. Using the power dissipation curves for a 5-V, 8-Q system, the internal dissipation in the TPA311
and maximum ambient temperatures is shown in Table 3.

Table 3. TPA311 Power Rating, 5-V, 8-0., BTL
MAXIMUM AMBIENT
TEMPERATURE

PEAK OUTPUT POWER
(mW)

AVERAGE OUTPUT
POWER

POWER
DISSIPATION
(mW)

350

350mW

600

46°C

114°C

350

175 mW (3 dB)

500

64°C

120°C

350

88 mW (6 dB)

380

85°C

125°C

350

44mW(9dB)

300

98°C

125°C

350

22 mW (12 dB)

200

115°C

125°C

350

11 mW(15dB)

180

119°C

125°C

o CFMSOIC

OCFMDGN

Table 3 shows that the TPA311 can be used to its full 350-mW rating without any heat sinking in still air up to

46°C.

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3-153

3-154

TPA701
700·mW MONO LOW·VOLTAGE AUDIO POWER AMPLIFIER
u OR iJGiIi PACKAGE

• Fully Specified for 3.3-V and 5-V Operation
• Wide Power Supply Compatibility
2.5V-5.5V

(TOP VIEW)

• Output Power for RL =8 n
- 700 mW at Voo = 5 V, BTL
- 250 mW at Voo 3.3 V, BTL
• Ultra-Low Quiescent Current in Shutdown
Mode ••. 1.5 nA
• Thermal and Short-Circuit Protection
• Surface-Mount Packaging
- SOIC
- PowerPADTM MSOP

SHUTDOWN
BYPASS

VoGND

IN+
IN-

Vo+

=

VDD

description
The TPA701 is a bridge-tied load (BTL) audiO power amplifier developed especially for low-voltage applications
where internal speakers are required. Operating with a 3.3-V supply, the TPA701 can deliver 250-mW of
continuous power into a BTL 8-n load at less than 0.6% THD+N throughout voice band frequencies. Although
this device is characterized out to 20 kHz, its operation was optimized for narrower band applications such as
wireless communications. The BTL configuration eliminates the need for external coupling capacitors on the
output in most applications, which is particularly important for small battery-powered equipment. This device
features a shutdown mode for power-sensitive applications with a supply current of 1.5 nA during shutdown.
The TPA701 is available in an 8-pin sOle surface-mount package and the surface-mount PowerPAD MSOP,
which reduces board space by 50% and height by 40%.

?

VOO 6
RF

.,L.

Audio

.Y

VOO/2

Input

~C

r

RI

I

4

IN-

3

IN+

2

BYPASS

CBT

-=

,
,

From System Control

...

~

1

Cs

~

r
, - -,
,
,
,
'vv,
,
,
,
,
L-~
,

SHUTOOWN

I

r

-

•

: Y

±

VOO

VO+ 5

-=

J

1

Biasi

L Control

Vrr

8]"(
-....

700mW

7

GNO

11-

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

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Copyright © 2000, Texas Instruments Incorporated

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TPA701
70D-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S229B - NOVEMBER1998 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

SMALL OUTLINEt
(D)

MSO~
(DGN)

-40°C to 85°C

TPA7010

TPA7010GN

MSOP
SYMBOLIZATION
ABA

.. to 350 mW; 700 mW
t In the SOIC package, the maximum RMS output power IS thermally IimHed
peaks can be driven, as long as the RMS value is less than 350 mW.
:j: The 0 and OGN packages are available taped and reeled. To order a taped and reeled part, add

the suffix R to the part number (e.g., TPA701 DR).

Terminal Functions
TERMINAL
NAME
BYPASS

NO.

110

DESCRIPTION

I

BYPASS is the tap to the voltage divider for intemal mid·supply bias. This terminal should be connected to
a O.I·I1F to 2.2·I1F capacitor when used as an audio amplifier.
IN- is the inverting input. IN- is typically used as the audio input terminal.

2

GNO

7

IN-

4

I

GNO is the ground connection.

IN+

3

I

IN + is the non inverting input. IN + is typically tied to the BYPASS terminal.

SHUTDOWN

1

I

SHUTDOWN places the entire device in shutdown mode when held high (100

VOO

6

VO+

5

Vo-

8

=1.5 nA).

VOO is the supply voltage terminal.

0
0

VO+ is the positive BTL output.
Vo- is the negative BTL output.

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§
Supply VOltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

TAS25°C

DERATING FACTOR

TA = 70°C

TA=85°C

0

725mW

5.8mWrC

464mW

377mW

OGN

2.14 w'II

17.1 mWrC

1.37W

1.11 W

11 Please see the Texas Instruments document. PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002). for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
Operating free-air temperature, TA

~TEXAS

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MIN

MAX

2.5

5.5

V

-40

85

°c

UNIT

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B - NOVEMBERl998 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, Voo =3.3 V, fA =25"C (uniess oiherwise
noted)
PARAMETER

TEST CONDITIONS

Voo

Output offset voltage (measured differentially)

See Note 1

PSRR

Power supply rejection ratio

VOO = 3.2 V to 3.4 V

100

Supply current

BTL mode

Supply current, shutdown mode (see Figure 4)

See Note 2

IOO(SD)
NOTES:

MIN

TYP

MAX

UNIT

20

mV

1.25

2.5

mA

1.5

1000

nA

85

dB

1. At 3 V < VOO < 5 V the dc output voltage is approximately Vool2.
2. This parameter is measured wHh no extemal capacitors connected to the device.

operating characteristics, Voo = 3.3 V, TA = 25°C, RL = 8 n
PARAMETER

TEST CONDITIONS

MIN

Po

Output power, see Note 3

THO = 0.2%,

See Figure 9

THO+N

Total harmonic distortion plus noise

PO=250mW,

f = 200 Hz to 4 kHz,

See Figure 7
See Figure 7

TYP

MAX

250

UNIT
mW

0.55%

BOM

Maximum output power bandwidth

Gain =2,

THO =2%,

Bl

Unity-gain bandwidth

Open Loop,

See Figure 15

20

kHz

1.4

MHz

Supply ripple rejection ratio

f= 1 kHz,

CB=lI1F,

See Figure 2

79

dB

Vn

Noise output voltage

Gain = 1,

CB=O.lI1F,

See Figure 19

17

I1V(rms)

NOTE 3: Output power is measured at the output terminals of the device at f = 1 kHz.

electrical characteristics at specified free-air temperature, Voo =5 V, TA =25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

VOO

Output offset vo~age (measured differentially)

PSRR

Power supply rejection ratio

100

Supply current

IOO(SOI

Supply current, shutdown mode (see Figure 4)

operating characteristics, Voo

MIN

TYP

MAX

mV

1.25

2.5

mA

5

1500

nA

78

VOO=4.9Vt05.1 V

UNIT

20

dB

= 5 V, TA = 25°C, RL = 8 n

PARAMETER

TEST CONDITIONS

MIN

TYP
700t

Po

Output power

THO =0.5%,

See Figure 13

THO+N

Total harmonic distortion plus noise

Po = 250 mW,

f = 200 Hz to 4 kHz,

See Figure 11
See Figure 11

MAX

UNIT
mW

0.5%

BOM

Maximum output power bandwidth

Gain =2,

THO =2%,

Bl

Unity-gain bandwidth

Open Loop,

See Figure 16

20

kHz

1.4

MHz

Supply ripple rejection ratio

f= 1 kHz,

CB=lI1F,

See Figure 2

80

dB

Vn

Noise output voltage

Gain = 1,

CB=O.lI1F,

See Figure 20

17

I1V(rms)

t The OGN package, properly mounted, can conduct 700 mW RMS power continuously. The 0 package, can only conduct 350 mW RMS power
continuously, with peaks to 700 mW.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75.265

3-157

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S2298- NOVEMBERl998 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

VDD 6

~

RF
Audio
Input

~c~

RI
-A

I

4

IN-

3

IN+

,"

2

BYPASS

,
,

-- -

,
,
,
,
,
,
,
,
,
,
,

CB-:::~

I

,
1

J, Cs

.
.

VD[)f2

SHUTDOWN

~

J

~

Y

,",

Bias
r Control

I
VO+ 5

'v-

RL=8
'v-

~

Vo- 8

y

I

7
GND

!l

Figure 1. BTL Mode Test Circuit

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Frequency

100

Supply current

vs Supply voltage

3,4

Output power

vs Supply voHage

Po

5

vs Load resistance
vs Frequency

6
7,8,11,12

vs Output power

9,10,13,14

THO+N

Total harmonic distortion plus noise
Open loop gain and phase

vs Frequency

15,16

Closed loop gain and phase

vs Frequency

17,18

Vn

Output noise voltage

vs Frequency

19,20

Po

Power dissipation

vs Output power

21,22

~TEXAS

3-156

2

Supply ripple rejection ratio

INSTRUMENTS
POST OFFICE BOX 855303 • DALLAS, TEXAS 75265

VDD

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B- NOVEMBER1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY

SUPPLY CURRENT
vs
SUPPLY VOLTAGE

0
III

1.8
RL=8n
CB=1I1F
BTL

-10

'a

I

I

c
.2

Ia:

.!!

-20

I

a.
a.

-70

f1I

-80

1.4

C
~

-50

-eo

:::I

'E"

-40

b

a:

1.6

-$

a.
a.

B~L

:::I

0

-aa.

./

,...--~ ~

1.2

---

~

:::I

f1I
'OICii

~DD=3.3V

"

-90

..",

VDD=5V

1111111

-100
20

100

I

'"

Q

E
0.8

I
10k

1k
f - Frequency - Hz

0.6

20k

2.5

3.5

3

4

4.5

5

5.5

VDD - Supply Voltage - V

Figure 2

Figure 3
SUPPLY CURRENT
vs
SUPPLY VOLTAGE

10

SHUTDOWN

9

=High

8

'"
c

7

I

I

6

0

5

:::I

4

-aa.

/

/"

f1I

I
Q

E

-

3

-

2

o

2.5

~

3

./

/

/'

3.5
4
4.5
VDD - Supply Voltage - V

5

5.5

Figure 4

~1ExAs

INSTRUMENTS
POST OFFICE

eox 655303 •

DALlAS. TEXAS 75265

3-159

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPUFIER
SLOS229B- NOVEMBER1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
SUPPLY VOLTAGE
1000 ....---,---,----r--.,.---..,----,
THD+Nl%
f= 1 kHz
BTL

800 1-----r---+----1--+---*----1

~r_-~-_r-___!-~+_-~-_;

VDD - Supply Voltage - V

Figure 5
OUTPUT POWER
vs
LOAD RESISTANCE
800
700

~I

600

I

500

5

400

0

300

0
Do

~
I

~

THD+N = 1%
f=l kHz
BTL

~

\

1\VDD=5V

" "-

~:,=3.3V
.......

200

100

o

8

16

.............

i'...

~

r--

-- -

24
32
40
48
RL - Load Resistance - n

-

56

Figure 6

~TEXAS

INSTRUMENTS
3-160

POST OFACE BOX 655303 • DAllAS, TEXAS 75265

64

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B - NOVEMBER1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10
~
I

vs

FREQUENCY

FREQUENCY
10

1=

~
I

VOO=3.3V
PO=250mW
t- RL=S(l
r- BTL

t=

.~

Z

+
c

IIII

~

II)
AV =-20 VN

I

I

AV=-10VN

Q

~

c

~

..Ai

I'"

0.1

t=I
Z

Ii

~

0.1

PO=125mW

li
t=I

.

+

Q

:r

j:

Po = 250 mW

I-

0.01
20

100

1k

10k

1 -I J I J 1111

0.01

20k

100

20

1k

f - Frequency - Hz

vs

OUTPUT POWER

OUTPUT POWER
10

:: VOO=3.3V

.~

==

'iP-

~~~~~zVN

I

.~

1/

_ BTL

+
c

Z
+

/

~
~

c
0

.

E

-I-

RL=S(l

~

r--

c

~
:r

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

Z

'2

F;;;;;:. f = 1 kHz

t=I

Z
+
Q

~

~0

Ili

li
t=I

f=20kHz

E=

~
"0

I

0.1

f=10kHz

i-0.1

1===
f= 20 Hz
t--

Voo = 3.3 V
RL=S(l
CB = 1 J.lF
AV =-2 VN
BTL
I I I LLL

Z
+

Q

:r

:r

I-

0.01

I-

o

0.05

0.1

20k

Figure 8

TOTAL HARMONIC DISTORTION PLUS NOISE

I

10k

f - Frequency - Hz

Figure 7

~

=

-,

Z

r-

+

/.V'

0

:r

'=

IV.

c

AV =-2 VN

!L

Q

Po=--?OmW

~
.2

1/ V

I-"

-

0

VOO = 3.3 V
RL=S(l
_ AV =-2 VN
_ BTL

+
c

/

~

II

1=

f:

I

1/1"
~

S

.!!!

Ili

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

0.15

0.2

0.25

0.3

0.35

0.4

0.01
0.01

Po - Output Power - W

0.1
Po - Output Power - W

Figure 9

Figure 10

~TEXAS

INSTRUMENTS
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3-161

TPA701
70o-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S229B - NOVEMBER1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

'#.

10

VOO=5V
Po =700 mW
RL=80
BTL

I

J
+

'#.

-

c

Voo=5V
RL=80
t- Av=-2VN
I- BTL

!z

V/

AV=-20~,It,/

V

~

1=

1=

I

+

Po=50mW;

~

c

~

s

~

is

r-

~

I

is
AV=-l0'V.!'

~~

f'

0.1

B

~

Ai"=-2 VN

!

i

0.1

~

I

PO=350mW

~

Z

0
:z:

...

0.01
20

100

lk

10k

0.01

20k

vs

OUTPUT POWER

OUTPUT POWER

~

VOO=5V
I- f=lkHz
I- AV=-2VN
I- BTL

'#.

,

i!

~

+
c

~

i

Q

i!
i

--

III

-5z

i

:z:

10

I

I

0

III

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

+
c

....t= 20 kHz
f= 10 kHz

c

0.1

'=1 kHz

0

..-l

!
!

RL= 8ri

I I Ii

~I

Z

Z

...:z:

i!:

0

+

Q

0.01
0.1

0.2

0.3

0.4 0.5

0.6

0.7

0.8

0.9

1

f=20Hz

0.1

i

~I

0.01
0.01

r-- t--

VOO=5V
RL=80
CB=l ~F
AV =-2 VN
BTL

Po - Output Power ... W

0.1
Po - Output Power - W

Figure 13

Figure 14

~TEXAS

3-162

r:::.~

.Ii

t--

20k

Figure 12

TOTAL HARMONIC DISTORTION PLUS 'NOISE

J

10k

f - Frequency - Hz

Figure 11

I

lk

100

20

f - Frequency - Hz

'#.

=

I

L

-

Z

0

:z:

[/

W

PO=700mW

i

~

...

~

.~

~

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B- NOVEMBER1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY

80

70
60
ID

50

I

40

1800

I"

"gc.
~

8.

0

1400
100

Phase

"

'Q

i

VOO=3.3V
RL=Open
BTL

30

r""--i'o

60 0

20

Gain

20

0

r--...

0

J

-200C.

I'

10

I'-..

0

i'o

_100 0

-10
-1400

-20
-30

104

1

-1SOO

f - Frequency - kHz

Figure 15
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY

80

70
60
ID

50

I

40

".9t
i

30

1800

"

0

20

140

0

1000

Phase

"

'Q

c
'ii

VOO=5V
RL=Open
BTL

....... ~

600

Gal~

10

-6(10

"

0
-10

~

-100

0

-1400

-20
-30
1
f - Frequency - kHz

103

104

_180 0

Figure 16

~TEXAS

INSTRUMENTS
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3-163

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B- NOVEMBER1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE
VB
FREQUENCY

,,-

0.75
0.5
ID

0.25

I

0

~

j

-G.25

!

-G.5

i

S

/

180°
Phase .........

/

"\

I

/'

180"

""\

Gain

{

\
\
\

-G.75
-1
-1.25

-1.5
-1.75

170°

\

VDD=3.3V
RL=80
Po = 250 mW
BTL

,

102

103

104

J
II.

140°
~

r-----T---

-2
101

150°

1\
\

105

130°

106

120°

f - Frequency - Hz

Figure 17
CLOSED-LOOP GAIN AND PHASE
VB
FREQUENCY
. , , - Phase

0.75
0.5
ID

0.25

I

0

~

c

ii
CJ

Do

i

-G.25
-G.5

I

/

-"

180°

\

I

{

/

170°

-'

160°

""\'

Gain

\

150°

-G.75
140°

-1
-1.25
-1.5
-1.75

-2

101

VDD=5V
RL=80
PO=700mW
BTL

\
\

\ '

I
104
f - Frequency - Hz

Figure 18

~lEXAS

3-164

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

130°

120°

106

J

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B - NOVEMBER1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100

~

OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100

: VOO=3.3V
BW = 22 Hz to 22 kHz
RL=800r320
AV=-1 VN

Voo:::iS"V
BW = 22 Hz to 22 kHz
RL=800r320
AV=-1 VN

,
~

I

I

t

II

VOBTL

~

..
z

Vo+

~

:u..
Vo+

II

10

~

VOBTL

~

Ci

:s

10

:s

!
0

!

0

I

I

::f"

::f"
1
20

100

1k

10k

1
20

20k

1k

100

f - Frequency - Hz

Figure 19

vs

OUTPUT POWER

OUTPUT POWER

/'

-"""""'t'....

~

-............

c

200

is

I

o

V

I
C

Ia.

500

5

400

.I

-

RL=80 -

/

~

~

L

300

"
o

600

I lL/
I

~20

50

E

'iii

150
100

I

BTL Mode
VOO=5V

700

RL=80

300

/
i....
/
II
I,p

800

i

BTL Mode
VOO=3.3V

250

POWER DISSIPATION

vs
350

20k

Figure 20

POWER DISSIPATION

~I

10k

f - Frequency - Hz

Q

a.

"

200

L

100

200
400
Po - Output Power - mW

600

o

o

~20

~ 1'-0..
200

400

600

800

1000

Po - Output Power - mW

Figure 21

Figure 22

~TEXAS

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TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B - NOVEMBER1998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load
Figure 23 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA701 BTL amplifier consists
of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive
configuration but initially consider power to the load. The differential drive to the speaker means that as one side
is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the
load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where voltage is
squared, yields 4x the output power from the same supply rail and load impedance (see equation 1).
V

_ VO(PP)
(rms) 2/2

2
V(rms)
Power = - - -

(1 )

RL

Voo

J';
RL

J'!
'V;

vO{PP)

2x VO{PP)

-VO(PP)

Figure 23. Bridge-TIed Load Configuration
In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an

8-n speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is
a 6-dB improvement, which is loudness that can be heard. In addition to increased power, there are frequency
response concerns. Consider the single-supply SE configuration shown in Figure 24. A coupling capacitor is
required to block the dc offset voltage from reaching the load. These capacitors can be quite large
(approximately 33 J.LF to 1000 J.LF) so they tend to be expensive, heavy, occupy valuable PCB area, and have
the additional drawback of limiting lOW-frequency performance of the system. This frequency limiting effect is
due to the high pass filter network created with the speaker impedance and the coupling capacitance and is
calculated with equation 2.
(2)

~TEXAS

3-166

INSTRUMENTS
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TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B - NOVEMBER1998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load (continued)
For example, a 68-IlF capacitor with an 8-il speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

VOO

~dB~-----J~=====

Figure 24. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of a SE
configuration. Internal dissipation versus output power is discussed further in the thermal considerations
section.

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from Voo.
The internal voltage drop multiplied by the RMS value of the supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply
to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the
amplifier, the current and voltage waveform shapes must first be understood (see Figure 25).
'DO

/

V(LRMS)

-~-

'OO(RMS)

Figure 25. Voltage and Current Waveforms for BTL Amplifiers

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SL0S229B - NOVEMBER1998 - REVISED MARCH 2000

APPLICATION INFORMATION

BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
PL
Efficiency = - -

(3)

P SUP

Where:

= VOO IOOrms

2Vp
= 11: RL

Efficiency of a BTL Configuration =

(4)

VP
2VOO
11:

Table 1 employs equation 4 to calculate effiCiencies for three different output power levels. The efficiency of the
amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resulting in a
nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output
power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper
power supply design.
Table 1. Efficiency Vs Output Power in 3.3-V 8-0 BTL Systems
(W)

EFFICIENCY
(%)

PEAK-to-PEAK
VOLTAGE
(V)

INTERNAL
DISSIPATION

0.125

33.6

1.41

0.26

0.25

47.6

2.00

0.29

2.45t
58.3
0.375
t High-peak voltage values cause the THO to increase.

0.28

OUTPUT POWER

(W)

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. In equation 4, Voo is in the denominator. This indicates
that as Voo goes down, efficiency goes up.

~TEXAS

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INSTRUMENTS
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TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S229B - NOVEMBER1998 - REVISED MARCH 2000

APPLICATION INFORMATION

application schematic
Figure 26 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of
-10 VN.

Voo 6

RF
50kO
Audio
Input

~CI

r-~~----~--~~-------------+--~---'~---Voo

Vo1)/2

-=-

RI
10kO

4

IN-

3

IN+

2

BYPASS

T

Cs
111F

Vo+ 5

-=-

CB
2.211F

T
-=-

Vo- 8

700mW

7
GND
From System Control

1

SHUTDOWN

Figure 26. TPA701 Application Circuit
The following sections discuss the selection of the components used in Figure 26.

component selection
gain setting resistors, RF and RI
The gain for each audio input of the TPA701 is set by resistors RF and RI according to equation 5 for BTL mode.
BTL Gain

= -

2(~~)

(5)

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA701 is a MOS amplifier, the input impedance is very high;
consequently input leakage currents are not generally a concern, although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 k.Q and 20 k.O. The effective impedance is calculated in equation 6.
Effective Impedance = R

RR
~~
F

(6)

I

~TEXAS

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TPA701
700-mW MONO LOW-VOLT"GE AUDIO POWER AMPLIFIER
SL0S229B - NOVEMBER1999 - REVISED MARCH 2000

APPLICATION INFORMATION
component selection (continued)
As an example consider an input resistance of 10 kn and a feedback resistor of 50 k.O. The BTL gain of the
amplifier would be -1 0 VN and the effective impedance at the inverting terminal would be 8.3 kn, which is well
within the recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kn, the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance ofthe MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kn. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.

~dB~====~~-----(7)

fc(lowpass)

For example, if RF is 100 kn and CF is 5 pF, then feo is 318 kHz, which is well outside of the audio range.

, input capacitor, C,
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper de level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.

fC(highpass) =

2lt~ICI

(8)

The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kn and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
(9)

~TEXAS

INSTRUMENTS
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TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S229B - NOVEMBER199B - REVISED MARCH 2000

APPLICATION INFORMATION

component selection (continued)
In this example, CI is 0.40 IlF, so one would likely choose a value in the range of 0.47 IlF to 1 IlF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI' CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage atthe inputto the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications, as the dc level there is held at Vool2, which is likely higher
than the source dc level. It is important to confinn the capacitor polarity in the application.

power supply decoupllng, Cs
The TPA701 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also prevents
oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved
by using two capacitors of different types that target different types of noise on the power supply leads. For
higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 IlF placed as close as possible to the device Voo lead works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 IlF or greater placed near the audio
power amplifier is recommended.

midrall bypass capacitor, Ca
The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CB detennines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the mid rail generation circuit internal to the amplifier, which appears as degraded PSRR and
THO + N. The capacitor is fed from a 250-k.O source inside the amplifier. To keep the start-up pop as low as
possible, the relationship shown in equation 10 should be maintained. This insures the input capacitor is fully
charged before the bypass capacitor is fully charged and the amplifier starts up.
10
(C B x 250

<

1

kn) - (RF + RI) CI

(10)

As an example, consider a circuit where CB is 2.2IlF, CI is 0.47IlF, RF is 50 kO, and RI is 10 k.O. Inserting these
values into the equation 10 we get:
18.2:s 35.5
which satisfies the rule. Bypass capacitor, Ce, values of 0.11lF to 2.21lF ceramic ortantalum low-ESR capacitors
are recommended for the best THO and noise performance.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capaCitor in the circuit. The lower the equivalent value of this
resistance, the more the real capaCitor behaves like an ideal capacitor.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • OALLAS, TEXAS 75265

3-171

TPA701
70Q-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S229B- NOVEMBER1998 - REVISED MARCH 2000

APPLICATION INFORMATION

5-V versus 3.3-V operation
The TPA701 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability.
The most important consideration is that of output power. Each amplifier in TPA701 can produce a maximum
voltage swing of Voo - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) 2.3 V as
opposed to VO(PP) = 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into
~n 8-n load before distortion becomes significant.

=

Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power of operation from 5-V supplies for a given output-power level.

headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA701 data sheet, one can see that when the TPA701
is operating from a 5-V supply into a 8-n speaker that 700 mW peaks are available. Converting watts to dB:
PdB

=

P

10Log--..Yt.
P ref

= 10Log

700 mW
1W

= -1.5

dB

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
-1.5
-1.5
-1.5
-1.5
-1.5

=

dB -15 dB -16.5 (15 dB headroom)
dB -12 dB = -13.5 (12 dB headroom)
dB - 9 dB = -10.5 (9 dB headroom)
dB - 6 dB -7.5 (6 dB headroom)
dB - 3 dB = -4.5 (3 dB headroom)

=

Converting dB back into watts:
Pw = 10PdB/10 x P ref
= 22 mW (15 dB headroom)
= 44 mW (12 dB headroom)
= 88 mW (9 dB headroom)
= 175 mW (6 dB headroom)
= 350 mW (3 dB headroom)

~TEXAS

INSTRUMENTS
3-172

POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

TPA701
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS229B- NOVEMBER1998 - REVISED MARCH 2000

APPLICATION INFORMATION

headroom and thermal considerations (continued)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with 0 dB
of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings
for the system. Using the power dissipation curves for a 5-V, 8-il system, the internal dissipation in the TPA701
and maximum ambient temperatures is shown in Table 2.
Table 2. TPA701 Power Rating, S-V, 8-0., BTL
PEAK OUTPUT
POWER
(mW)

AVERAGE OUTPUT
POWER

POWER
DISSIPATION
(mW)

DPACKAGE
(SOIC)

DGNPACKAGE
(MSOP)

MAXIMUM AMBIENT
TEMPERATURE

MAXIMUM AMBIENT
TEMPERATURE

1100 e

700

700mW

675

34°e

700

350 mW (3 dB)

595

47°e

115°e

700

176 mW (6 dB)

475

68°e

122°e

700

88 mW(9 dB)

350

700

44 mW (12 dB)

225

89°e
moe

125°e

125°e

Table 2 shows that the TPA701 can be used to its full 700-mW rating without any heat sinking in still air up to
110°C and 34°C for the DGN package (MSOP) and D pacakge (SOIC) respectively.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-173

3-174

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
1998 - REViseD MARCH 2000

• Fully Specified for 3.3-V and 5-V Operation
• Wide Power Supply Compatibility
2.5V-5.5V

• BTL to SE Mode Control
• Integrated Depop Circuitry
• Thermal and Short-Circuit Protection

• Output Power
- 700 mW at Voo = 5 V, BTL, RL = 8 0
- 85 mW at Voo 5 V, SE, RL 32 0
- 250 mW at Voo 3.3 V, BTL, RL 8 0
- 37 mWat Voo = 3.3 V, SE, RL = 32 0

• Surface-Mount Packaging
- SOIC
- PowerPADTM MSOP

=
=

=

=

o OR OGN PACKAGE
(TOP VIEW)

• Shutdown Control
- 100 7 IlA at 3.3 V
- 100 50 IlA at 5 V

=
=

Vo-

SHUTDOWN
BYPASS
SElBTL
IN

description

GND
VDD

Vo+

The TPA711 is a bridge-tied load (BTL) or
single-ended (SE) audio power amplifier developed especially for low-voltage applicationswhere internal speakers and external earphone operation are
required. Operating with a 3.3-V supply, the TPA711 can deliver 250-mW of continuous power into a BTL 8-0
load at less than 0.6% THD+N throughout voice band frequencies. Although this device is characterized 'out
to 20 kHz, its operation was optimized for narrower band applications such as wireless communications. The
BTL configuration eliminates the need for external coupling capacitors on the output in most applications, which
is particularly important for small battery-powered equipment. A unique feature of the TPA711 is that it allows
the amplifier to switch from BTL to SE on the flywhen an earphone drive is required. This eliminates complicated
mechanical switching or auxiliary devices just to drive the external load. This device features a shutdown mode
for power-sensitive applications with special depop circuitry to eliminate speaker noise when exiting shutdown
mode. The TPA711 is available in an 8-pin SOIC and the surface-mount PowerPAD MSOP package, which
reduces board space by 50% and height by 40%.

VOO 6
VOO

RF

~CI

Voot2

-=-

Audio
Input
R,

4

IN

2

BYPASS

r
VO+ 5

CBr

CS

-=-

~
-=-

-=-

700mW

Vo- 8
7
From System Control
From HPJack

•

~

1

GND

SHUTDOWN

-=-

3 SElBTL

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3-175

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS23OB- NOVEMBER 1998 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

SMALL OUTLINEt
(D)

MSO~

TPA7110

TPA7110GN

-4O"C to 85°C

MSOP
SYMBOLIZATION

(DON)

ABB

t In the SOIC package, the maximum RMS output power Is thermally limited to 350 mW; 700 mW
peaks can be driven, as long as the RMS value Is less than 350 mW.
:I: The 0 and OGN packages are available taped and reeled. To order a taped and reeled part, add
the suffix R to the part number (e.g., TPA3110R).

Terminal Functions
TERMINAL
NAME

NO.

110

DESCRIPTION

I

BYPASS is the tap to the voltage divider for intemal mid·supply bias. This terminal should be connected to
a 0.1-I1F to 2.2-I1F capacitor when used as an audio amplifier.

BYPASS

2

GNO

7

IN

4

I

IN is the audio input terminal.

SElBTL

3

I

When SElBTL is held low,the TPA711 is in BTL mode. When SElBTL is held high, the TPA711 is in SE mode.

SHUTDOWN

1\

I

VOO

6

VO+

5

Vo-

8

GNO Is the ground connection.

SHUTDOWN placas the entire device In shutdown mode when held high (100

=71lA).

VOO is the supply voltage terminal.

0
0

Vo+ is the positive output for BTL and SE modes.
Vo- is the negative output in BTL mode and a high-impedance output in SE mode.

absolute maximum ratings over operating free-alr temperature range (unless otherwise noted)§
Supply voltage, Voo ........................................................................ 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maximum ratings" may cause pemianent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

TAS25°C

DERATING FACTOR

0

725mW

5.8mW/"C

=

TA 70"C
4B4mW

=

TA 85°C
377mW

OGN
2.14 w1I
17.1 mW/"C
1.37W
1.11 W
11 Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMAOO2), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
Operating free-air temperature, TA (see Table 3)

~TEXAS

3-176

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

MiN

MAX

2.5

5.5

UNIT
V

-40

85

°C

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

=

=

electrical characteristics at specified free-air temperature, VDD 3.3 V, TA 25°C (unless otherwise
noted)
PARAMETER
Voo

TEST CONDmONS

Output offset voltage (measured differentially)

MIN

Power supply rejection ratio

100

Supply current (see Figure 6)

IOO(SO)

Supply current, shutdown mode (see Figure 7)

MAX
20

I BTL mode

PSRR

TYP

See Note 1
VOO = 3.2 V to 3.4 V

85

I SE mode

UNIT
mV
dB

83

BTL mode

1.25

2.5

SEmode

0.65

1.25

7

50

TYP

MAX

mA

IIA

NOTE 1: At 3 V < VDD < 5 V the de output voltage is approximately VDoI2.

operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 8 n
PARAMETER

TEST CONDITIONS

MIN

THD = 0.2%,

BTL mode,

See Figure 14

THD=O.l%,
See Figure 22

SEmode,

RL=32n,

Po

Output power, see Note 2

THD+N

Total harmonic distortion plus noise

Po=250mW,

1= 200 Hz to 4 kHz,

See Figure 12

BOM

Maximum output power bandwidth

Gain=2,

THD = 2%,

See Figure 12

B1

Unity-gain bandWidth

Open Loop,

See Figure 36

1=1 kHz,
See Figure 5

CB=l).1F,

BTL mode,

1= 1 kHz,
See Figure 3

CB=l).1F,

SEmode,

Gain = 1,

CB=O.l ).1F,

See Figure 42

Supply ripple rejection ratio

Vn

Noise output voltage

NOTE 2: Output power is measured at the output terminals 01 the device at 1 = 1 kHz.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

UNIT

250
37

mW

0.55%
20

kHz

1.4

MHz

79
dB
70
17

).1V(rms)

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, Voo
noted)

TEST CONDmoNS

PARAMETER
VOO

=5 V, TA =25°C (unless otherwise
TYP

Power supply rejection ratio

IDD

Supply current (see Figure 6)

IDDISDI

Supply current, shutdown mode (see Figure 7)

operating characteristics, Voo

VDD = 4.9 V to 5.1 V

Output power, see Note 2

THD+N

Total harmonic distortion plus
noise

I SE mode

BTL mode

1.25

2.5

0.65

1.25

50

100

TEST CONDITIONS

MIN

THD = 0.3%,

BTL mode,

See Figure 18

THD=O.l%,
See Figure 26

SEmode,

RL=32Q,

PO=700mW,

1 = 200 Hz to 4 kHz,

See Figure 16
See Figure 16

BaM

Maximum output power bandwidth

Gain =2,

THD=2%,

Unity-gain bandwidth

Open Loop,

See Figure 37

1= 1 kHz,
See Figure 5

CB= ll1F,

BTL mode,

1= 1 kHz,
See Figure 4

CB=lI1F,

SEmode,

Gain = 1,

CB=O.lI1F,

See Figure 43

Noise output voltage

mV
dB

76

SEmode

Bl

Supply ripple rejection ratio

78

UNIT

mA

jiA

= 5 V, TA = 25°C, RL = 8 n

PARAMETER
Po

MAX
20

I BTL mode

PSRR

Vn

MIN

Output offset voltage (measured differentially)

TYP

MAX

UNIT

700t

85

mW

0.5%
20

kHz

1.4

MHz

80
dB
73

17

I1V(rms)

t The DGN package, properly mounted, can conduct 700 mW RMS power continuously. The 0 package, can only conduct 350 mW RMS power
continuously, with peaks to 700 mW.
NOTE 2: Output power is measured at the output terminals 01 the device at 1 = 1 kHz.

~TEXAS

INSTRUMENTS
3-178

POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

l-

RF
Audio
Input

~C

RI

L

4

VOO 6

..A. A

-.

IN

I

2

BYPASS

.lT-=

J-

~

T-=

VO+ 5

VOO
Cs

r
, -, ~V

I
CB

-l.

VOot2

,
,
,
,
,
,
,
,
,
,
,
,
1

SHUTOOWN I

3

SElBTL

RL=8 n

r

-

--+--

I Control
Bias

~

IV
•

Vo- 8

7
GNO

r1-

JFigure 1. BTL Mode Test Circuit

VOO 6
VOO

RF
VOot2

-=

Audio
Input
RI

~CI
-=

4

IN

2

BYPASS

VO+ 5

CB

T-=
Vo- 8

T-=

l

Cs

RL=32n

-=

7
1

-=

SHUTOOWN ...-......_..,

GNO

3 SElBTL
VOO -=-1-=::..::..::'----1

L-_--J

Figure 2. SE Mode Test Circuit

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAI.lAS, TEXAS 75265

3-179

TPA711
70o-mW MONO LOW-VOLTAGE AUDIO POWER AMPLlFJER
SLOS230B- NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE

100

Supply ripple rejection ratio

vs Frequency

Supply current

vs Supply voltage

6,7

vs Supply voltage

8,9

Output power

Po

THO+N

vs Load resistance

III
'a

I

-10
-20

.2

ir:

-30

vs Frequency
vs Output power

14,15,18,19,22,23,
26, 27, 30, 31, 34, 35

Total harmonic distortion plus noise

Open loop gain and phase

vs Frequency

36,37

Closed loop gain and phase

vs Frequency

38,39,40,41

Vn

OUtput noise voltage

,vs Frequency

Po

Power dissipation

vs Output power

0

i

-40

l

~

.!I!

I

2'

a.
a.

~

,

vs
FREQUENCY

~
c

0

CB = 1'>-

"

-70

-80
-100

III
'a
0

'CB=0.1I1F

~=}I2IVIIII
II I
100

!O

"

""

"
I'\.

,

11'ii'
II:

-10

~

-40

",

-50

~

a.
a.
:::s

-70

III

I.:

-80

/
~
1/

BYPASS = 112 VDD

-100
20

f - Frequency - Hz

I'"100

'I ""1k

f - Frequency - Hz

Figure 4

Figure 3

~TEXAS

3-180

, ""

t-i"'"

-80
10k 20k

...........

CB=1I1F'
-60

2'

,

~B=0.1I1F

t-..

-30

,
VDD=5V
RL=80 SE

-20

CD

ii

i'J~

1k

,

0

I

~

-80 ;;;;;

SUPPLY RIPPLE REJECTION RATIO

FREQUENCY

I'.
-

42,43
44, 45, 46, 47

vs
VDD=3.3V
RL=80
SE

,

10,11
12,13,16,17,20,21,
24, 25, 28, 29, 32, 33

SUPPLY RIPPLE REJECTION RATIO

0

3,4,5

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B- NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO

SUPPLY CURRENT

vs
FREQUENCY

SUPPLY VOLTAGE

vs

0
RL=80"
CB =111F
BTL

-10

III

"D
I

J

-20

1.6
1.4

-30

1.2

V- ....-

0.8

/

-

BTL

SE

0.6
1"-1

,i,lif

-90
-100
20

l..".o- ~

~OO=3.3V

....
100

0.4
0.2

V
,

o

1k
f - Frequency - Hz

2.5

10k 20k

3

3.5

4

4.5

5

5.5

VOD - Supply Voltage - V

Figure 5

Figure 6
SUPPLY CURRENT

vs
SUPPLY VOLTAGE
90
SHUTDOWN

=High

80

cc::I.

70

I

60

iu

;:,

50

~
D.
D.
;:,

40

I

30

UJ

/

C

E

20
10

o

-

2.5

l--"
3

3.5

/

/

/

7

/

J

4

4.5

5

5.5

VOD - Supply Voltage - V

Figure 7

~TEXAS

INSTRUMENTS
POST OFFICE BOX 855303 • DAllAS. TEXAS 75265

3-181

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUT'PUT POWER

vs

vs

SUPPLY VOLTAGE

SUPPLY VOLTAGE

1~r---~---r---.----.---.-----,

350

THO+N1%
f=1 kHz
BTL

300

~

~I

I

I

600~--~---+--~--~+---~--~

J

.
I

250

~

150

_I

t-

100
50

o

~

Figure 9

OUTPUT POWER

OUTPUT POWER

I

,p

LOAD RESISTANCE
350

THO+N=1%
f=1kHz
BTL

\

~I

""",

0
II.

t

'"

300 ROO = 3.3 V

r-..

200
100

8

16

-...

r---..

24

i

["0...

--

32

56

200

~

~
I"\0O=5V

150

I

,p

48

250

f'.. t--...

0

r----- t--

40

THO+N=1%
f= 1 kHz
SE

300

I\.VOO=5V

400

o

100

l\

50

......

\'..t--.....

r---..... :---

---

r-- r--

VOO = 3.3 V
64

o

8

1

I

14

20

RL - Load Resistance - 0

Figure 10

I

26

32

INSTRUMENTS
POST OFRCE BOX 655303 • DALLAS, TEXAS 75265

38

44

50

RL - Load Resistance - 0

Figure 11

~TEXAS

3-182

5.5

5

vs

'\
700

!0

4.5

Figure 8

800

'S

4

~

VOO - Supply Voltage - V

LOAD RESISTANCE

500

~

I--'""'"

~

3.5

3

vs

I

~

RL=320

~ f..---

2.5

600

/

/

V

VOO - Supply Voltage - V

~I

/

./

I

,p

V

RL=80 /

0

I

/

200

'S

,p

I

THO+N=1%
'=1 kHz
SE

56

62

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

.,.

FREQUENCY

.,.
Iz

10
t- VOO=3.3V

I

:Il
"0
z

';::: PO=250mW
t- RL=SO
BTL

V

c

r--

/

+

~0

11
is

~0

VOO=3.3V
RL=SO
_ AV=-2VN
_ BTL

+

~
F

c

~0

Po =,SOmW

i5

1/1.'

I-'

~

u

A

~

0.1

1=
1=

11

AV=-10,YJV

-

i

:z:

V/

Jill 1i;'l20VN ",,'

I-~

10

I

'2

/X

0

i

Av=-2VN

:z:

j
~
I

~

0.1

PO=125mW

j
~I

Z

+

Z

C

~

C

..,

--=

+

...:z:

0.01

100

20

1k

10k

20k

Po = 250 mW
0.01

100

20

Figure 13

TOTAL HARMONIC DISTORTION PLUS NOISE

10

I

vs

OUTPUT POWER

OUTPUT POWER

.,.

~ VOO=3.3V

L
,

i

,g

c

:z:

RL=SO

r-

I

1/

~

~
01

I
j

~

'z7
o

I

Z
+
C

:z:

i!:

o

0.05

0.1

0.15

0.2

0.25

0.3

E

i

I

0.1

0.01

~

i!

0.35

0.4

-

f=2OkHz

~

+

c
o

j

...

10

I

c

0

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

~ f=1 kHz
t- AV=-2VN
t- BTL

+

20k

f - Frequency - Hz

Figure 12

.,.
Iz

10k

1k

f - Frequency - Hz

f=10kHz

~f=1kHz

t-.

0.1

F===
r---

0.01
0.01

Po - Output Power - W

f=20Hz

VOO=3.3V
RL=SO
CB= 111F
AV =-2 VN
BTL

0.1

I

I

""

Po - Output Power - W

Figure 14

Figure 15

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAL....S. TEXAS 75265

3-183

TPA711
70D-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS23OB- NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

10

il-

10

VOO=5V
Po =700 mW
RL=80
BTL

I

•

i

il-

J

V~

+

-

IS

AV=-20~.•/

:i

1
x:

0.1

I

~

~

oS!

c0

D l/

~

A~=-2VN

x:

0.1

J

~

PO=350mW -

Z

+

Q

0.01

100

1k

10k

0.01

20k

100

20

f - Frequency - Hz

I

rrr-

10

t=: VOO=5V

+

...

I

I

I
I

RL=80-1

0.1

...

i

J
~

~
I

~

~

r--

+

I

I
I

il-

,

f=1 kHz
AV=-2VN
Bn

to--

0.2

0.3

OA 0.5 0.6

0.7

0.8

0.9

1

r- t-

f=2OkHz

f=1 kHz
I I IT
f=20Hz

r-- r-

0.1

0.01
0.01

VOO=5V
RL=80
CB=1I1F
AV =-2 VN
Bn

Po - OUtput Power - W

0.1
Po - Output Power - W

Figure 19

Figure 18

~TEXAS

3-184

.... t-

f=10kHz

I

0.01
0.1

20k

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

10
l-

10k

Figure 17

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
I

1k
f - Frequency - Hz

Figure 16

il-

~

Po =700 mW

I

~
20

~

0

~

-

Z

0

Po=50mW -

c

A

~~

J

AV=-2VN
BTL

i

Av=-10y"'N

r--.

r-

;:

is

r-

VOO=5V

r-

+

V

i'Ii

1=

1= RL=80

I

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

.,.

vs

FREQUENCY

FREQUENCY

.,.

10
Voo = 3.3 V
PO=30mW
RL=320
SE

I

I
+

/":

i
0.1

J

i

'"

~

0.01

F AV=~VN

I

+

Po=10mW

0.1

~

0.01

I

i

1IIIlIi
20

I

J~

AV=-1 VN

III

0.001

f=

VOO=3.3V
~ RL=320
_ Av=-1 VN
SE

II

, / ,/

is

!~

10

I

AV=-10VN

.!!

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

100

1k

10k

20k

Po=15mW
pO=30mw'llill
I I

r0.001

11111
20

II 1111

100

f - Frequency -

Figure 20

vs

OUTPUT POWER

OUTPUT POWER

.,.

~ Voo=3.3V
t- f=1kHz
t- RL=320
t- Av=-1 VN
SE

I

~

+
c

~

/

1/

10

I

t=

I

t-

VOO=3.3V

~ RL=320

t- Av=-1 VN

+
c

SE

~

~

~
.!!

I

~

~
:z:

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

I

I

f=10kHz

0

Ii!III

:z:

!

J

~
I

~

Z

Z

I

+
Q

+
Q

:z:
~

0.01
0.02

f=20kHz

c

I

0.1

I

0.1

Kf=2OHz

r--

1=1 kHz

j!:
0.025

0.03

0.035

20k

Hz

Figure 21

TOTAL HARMONIC DISTORTION PLUS NOISE

.,.

10k

1k

f - Frequency - Hz

0.04

0.045

0.05

0.01
0.002

Po - Output Power - W

I
0.01

0.1

Po - Output Power - W

Figure 22

Figure 23

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75286

3-185

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

va

va

FREQUENCY

FREQUENCY

10

"l-

VOO=5V
PO=60mW
RL=320

I

Iz

fli

SE

+
c

~0
VOO=5V
1
fE:rtm!lm
RL=320
I- AV = -1 VN +-+-++t14-1+-----1H-+-I-t+f*--I

1=

f--

AV=-10V

r--

I

SE

AV=-5VN

~

~
~0

......

0.1

I!

i,;II

'"

"""

I::"§.::

....

0.01

~

./

~ ......

~

AV=-1VN ~

I

Z

+

Q

....:c

I II

0.001
100

20

10k

1k

20k

f - Frequency - Hz

f - Frequency - Hz

Figure 24

Figure 25

TOTAL HARMONIC DISTORnON PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

va

va

OUTPUT POWER

"l-

OUTPUT POWER

10

"l-

VOO=5V
f=1 kHz
RL=320
AV=-1 VN

I

I
+

.110

I

+

~

Col

0

E

!

~
I

Z

+
Q

:c
....

IIIII

~

_f :2OkHz

!1\1

0.1

:c

!

-

0.01
0.02

1kHz

0.1

~
I

I

-

Z

i!i

i!:

0.04

0.06

0.08

0.1

0.12

0.14

0.01
0.002

Po - Output Power - W

1=20 ~
kHz

III
0.01

Po - Output Power - W

Figure 27

Figure 26

~1ExAs

3-186

- ....

r--

~0

C

11

c

~

'f

Ii
:c

i= VOO=5V
1= RL=320

I- AV=-1 VN
I- SE

z

I

SE

IS

10

I

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

0.1

0.2

TPA711
70Q..mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B- NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

".

1=

I

vs

FREQUENCY

FREQUENCY

".

YOO=3.3Y

~ Po =0.1 mW

~Z
15

J

I

.!z

+

c

~

0.1
Ay=~YN

/'

0.01

.~

~

~

20

100

:!

0.01

,

Z
+
C

Po = 0.1 mW

1k

10k

0.001

20k

20

100

+
c

vs

OUTPUT POWER

OUTPUT POWER

1=

Ii
~I

YOO=3.3 Y
RL=10kn
Ay=-1 YN
SE

I

I

~

+

C

I

I

c

~0

10

".

§

i

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

YOO =3.3Y
~f=1kHz
I- RL=10kn
Ay=-1 YN
SE

!z

0.1

f= 20 Hz
0.1

.2

J
i~

0.01

f=20kHz

,.....

I

0.01

I

Z
+
C

~

:z:
I-

0.001
50

75

100

125

20k

Figure 29

TOTAL HARMONIC DISTORTION PLUS NOISE

10

10k

1k
f - Frequency - Hz

Figure 28

I

L

/

:z:

I-

f - Frequency - Hz

".

11111
Po = 0.05 mW "-

I

III
Ay=-1 YN

III JJ II

0

I§

5

Po = 0.13 mW

~

~ Ay=-2YN

~ 0.001

~~

0.1

i

i

'z7

YOO=3.3Y
RL=10kn
CS=1 J.1F
Ay=-1 YN
SE

I

- RL=10kn
:-- SE

+

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

150

175

200

f= 10kHz

f=1 kHz ~

0.001

5

Po - Output Power - J.1W

10

100

~~

500

Po - Output Power - J.1W

Figure 30

Figure 31

-!I1TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-187

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

I

FREQUENCY

z0

SE

SE

+
c

'"

0.1

VDD=5V
RL = 10 kn
Av=-1 VN

.;

t- RL=10kn

+

I
IJ

~
I

~ PO=0.3mW

J

0

'E

AV=..!J.V

.~

~

'Y

{!.
I

/

Z

PO=0.1 mW

0

:c

AV=-1 VN
100

20

10k

0.001

20k

100

20

f=

c

~
c

vs

OUTPUT POWER

OUTPUT POWER

10

+

~
.2

~

{!.

VDD=5V
RL = 10 kn
Av=-1 VN

iz

I

SE

+

SE

c

~

~..

0.1

·c0

f=20Hz

0.1

i

f=20kHz

:c

!

0.01

{!.

I

1'--

0.01

.L

Z

0

:c

f=10kHz

I-

0.001
50 100

150 200 250 300

350 400 450 500

I I I III

0.001

5

Po - Output Power - JlW

Figure 34

10

INSTRUMENTS
POST OFACE BOX 655303 • DALLAS, TEXAS 75265

r---

100
Po - Output Power - JlW

Figure 35

'!I1TEXAS

3-188

'"'"""

I

Z

0
i!:

10

I

0

I!

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

VDD=5V
- f=1 kHz
~ RL= 10kn
~ Av=-1 VN

20k

Figure 33

TOTAL HARMONIC DISTORTION PLUS NOISE

·1z

1k
f - Frequency - Hz

Figure 32

I

IIIIII10k

I-

1k
f - Frequency - Hz

~

I&~

!

N1 l.JM
0.001

~

Po=0.2mW

I~

:c 0.01

~~

1,\

I......

~

E
01

~

f:=

Po=0.3mW

0

~Ai~~~

0.01

~~

0.1

~

/

I

i!:

vs

FREQUENCY

1= VDD=5V

~

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

f= 1 kHz
I

I

I

500

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY

80
70

60
aI

50

I

c

40

CJ

30

180'

"

a.

g
....I

c!:

&

0

140'

Phase

100'

~

'Q

~

Voo = 3.3 V
RL = Open
BTL

I""-...r-.

60'
20'

Gain

20

r--....

10

J

-20'Q.

i'

-60'

I'...

0

r-.

-10

-100'
-140'

-20
-30

104

1

-180'

f - Frequency - kHz

Figure 36
OPEN-LOOP GAIN AND PHASE

'vs
FREQUENCY

80
70

60
aI

50

c

I

40

CJ

30

180'

"

a.
0

.9
c!:
!.
0

20

140'
100'

Phase

"

'Q

~

VOO=5V
RL=Opan
BTL

r--~

Gai~

10

60'

~

-60'

I'...

0

~

-100'

-10

_140'

-20
-30
1

102
f - Frequency - kHz

103

104

_180'

Figure 37

-!II
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-189

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SlOS23OB - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

V'

0.75
0.5
ID

"c
!
I

I

I

0.25

Phase-" ~

/

\

\
\
,\
\

I

0

/"

-0.25

Gain

{

-0.5

\

-0.75

\

-1
-1.25
-1.5
-1.75

VOO=3.3V
RL=8n
Po = 250 mW
BTL

-2
101

\

102

103

'\
\ \

104

105

r - Frequency - Hz

Figure 38
CLOSED-LOOP GAIN AND PHASE

vs

-

FREQUENCY
0.75
0.5
ID

"I
i
CI

a.

j

u

0.25
0
-0.25
-0.5

I

/

V

Phasa

180"

~

\

I

{

170"

\

/

160"

"\

Gain

\

\

150"

\

-0.75

140"

-1
-1.25

-1.5
-1.75

-2

101

1\

1\
\,

VOO=5V
RL=8n
PO=700mW
BTL

102

104
r - Frequency - Hz

Figure 39

~TEXAS

3-190

INSTRUMENTS
POST OFFlCE BOX 656303 • DALlAS. TEXAS 75265

130"

120"

106

I

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

180·

7

/

6

ID

"I
iCJ

.9

2

0

1

0
-1

-2
101

,

-

160·

-

150·

I

-

140-

I

-

130·

-

120·

~

\
Voo = 3.3 V
RL=320
AV=2VN
PO=30mW

1/

(j

170·

"'\ "\

/
II

4
3

Gain

/7

5

a.

io""'" Phase I"""

,

=

if

110·

SE

I

I

104

105

\

106

100·

f - Frequency - Hz

Figure 40
CLOSED·LOOP GAIN AND PHASE

vs
FREQUENCY
7

/

6
ID

"c

4

CJ

a.

3

10

2

I

"'\

I

0
0

"\ ,

~

I
II

iii

I

fl

0
-1

-2
101

Gain

/7

5

180·

~Phase~

VOO=5V
RL=320
AV=2VN
Po=60mW
SE

•

I

103

170·

-

160·

-

150·

-

140·

-

130·

-

120·

J

110·

\

106

1000

f - Frequancy - Hz

Figure 41

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

3-191

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS23OB- NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100

OUTPUT NOISE VOLTAGE
vs
FREQUENCY
100

1= VOO=UV

t- VOO=5V

~ BW=22Hzto22 kHz
t- RL=8nor32n
t-AV=l

'ii'

!

~BW=22Hzt022kHZ

'ii'

!

~

&

&

~
~

Vo.

10

z~

i

l'S

II
IJOBTL

I

VOBTL

~

J

RL=8nor32n
AV=1

~

I

~

t-

111111
Vo+

10

'S

t

0

0

I

I

~

~
1
20

100

1k

10k

1
20

20k

100

Figure 43

POWER DISSIPAnON
vs
OUTPUT POWER

POWER DISSIPATION
vs
OUTPUT POWER

350

100
90

300

I

~I

250

c

c
0

i

~
I
:.

80

RL=8n

70

I

60

I,p

40

/

I

I

,p
I-----~:-t------+ VOD = 3.3 V
BTL
0

400
Po - Output Power - mW
200

50

--...........

-.....

1

30

,.......RL=32n

20

,

o

o

Figure 44

""

~

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

VDD = 3.3 V
SE

100
50
Po - Output Power - W

Figure 45

~1ExAs

3-192

-

I'

10

600

--

L

Q

0

20k

f - Frequency - Hz

Figure 42

~

10 k

1k

f - Frequency - Hz

150

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS
POWER DISSIPATION
vs
OUTPUT POWER

POWER DISSIPATION
vs
OUTPUT POWER

800

200

700
~

E

600

0

I

500

300

200

100

o

0

I
I
S

I

RL=320

200

400

120

a.

VDD=5V

'"

BTL

60

800

~

/

RL=320

~

40

(

t---...

20

I

I'....

600

/
I

100
80

1000

o

--~

L

140

Q

~-.......

o

~I

RL=80

160

c

I
/

400

I 1
E
I

r---

,IV

I

c

180

RJ.=1 80

/ ' I---

o

PD - Output Power - mW

50

.........

1'..

100

VDD=5V
SE

........

~
150

1
200

250

300

PD - Output Power - mW

Figure 46

Figure 47

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-193

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load versus single-ended mode
Figure 48 shows a linear audio power amplifier (APA) in a BTL cOllfiguration. The TPA711 BTL amplifier consists
of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive
configuration but initially consider power to the load. The differential drive to the speaker means that as one side
is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the
load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where voltage is
squared, yields 4x the output power from the same supply rail and load impedance (see equation 1).
V

_ VO(PP)
(rms) 2/2

Power -

V(rms)

(1)

2

-~

Voo

J' :

RL

J'!
'V:

vO(PP)

2x VO(PP)

-VO(PP)

Figure 48. Bridge-Tied Load Configuration
In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an

8-n speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is
a 6-dB improvement, which is loudness that can be heard. In addition to increased power there are frequency
response concerns. Consider the single-supply SE configuration shown in Figure 49. A coupling capacitor is
required to block the dc offset voltage from reaching the load. These capaCitors can be quite large
(approximately 33 j.lF to 1000 j.lF) so they tend to be expensive, heavy, occupy valuable PCB area, and have
the additional drawback of limiting low-frequency performance of the system. This frequency limiting effect is
due to the high pass filter network created with the speaker impedance and the coupling capacitance and is
calculated with equation 2.
(2)

~TEXAS

INSTRUMENTS
3--194

POST OFACE BOX 655303 • DALlAS. TEXAS 75265

TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load versus single-ended mode (continued)
For example, a 68-IlF capacitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

Voo

~dB~----~~====

fe

Figure 49. Single-Ended Configuration and Frequericy Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Intemal dissipation versus output power is discussed further in the thermal considerations
section.

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from Voo.
The internal voltage drop multiplied by the RMS value of the supply current, IOorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply
to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the
amplifier, the current and voltage waveform shapes must first be understood (see Figure 50).
100

,/

V(LRMS)

--fVtfVVffll-

IOD(RMS)

Figure 50. Voltage and Current Waveforms for BTL Amplifiers

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TPA711
70D-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SlOS23OB- NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most ofthe waveform, both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

P
Efficiency = ~

(3)

SUP

Where:

Efficiency of a BTL Configuration

=

ltV

(4)

2V P

DO

Table 1 employs equation 4 to calculate efficiencies for three different output power levels. The efficiency of the
amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resulting in a
nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output
power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper
power supply design.
Table 1. Efficiency Vs Output Power In 3.3-V 8-0 BTL Systems

t

OUTPUT POWER

EFFICIENCY

(W)

(%)

0.125

33.6

PEAK-to-PEAK
VOLTAGE
(V)

INTERNAL
DISSIPATION

1.41

0.26

47.6
2.00
0.25
2.45t
58.3
0.375
High-peak voltage values cause the THO to Increase.

(W)

0.29
0.28

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. In equation 4, Vee is in the denominator. This indicates
that as Vee goes down, efficiency goes up.

~TEXAS

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TPA711
70D-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER. 1998 - REVISED MARCH 2000

APPLICATION INFORMATiON

application schematic
Figure 51 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of

-10VN.

RF

CF
5pF

VDD 6

501<0

.-~~----~--~v--------------+--~----~~---VDD

VDoJ2

Audio

Input
~

'I'
-=-

RI
101<0

CI
0.47 1lF

CB
2.2JLf

4

IN

2

BYPASS

Vo+ 5

CC
330llF

r

Cs
11lF

T
Vcr 8
7

From System Control

1

3
0.1 JLf

T

SHUTDOWN ....-'--........,
Bias
SElBTL
Control

GND

100 1<0

VDD--~~'-__----------------------------------------------~
100kQ

Figure 51. TPA711 Application Circuit
The following sections discuss the selection of the components used in Figure 51.

component selection
gain setting resistors, RF and RI
The gain for each audio input of the TPA711 is set by resistors RF and RI according to equation 5 for BTL mode.
BTL Gain

=-

2(~)

(5)

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA711 is a MOS amplifier, the input impedance is very high;
consequently input leakage currents are not generally a concern, although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kn and 20 kn. The effective impedance is calculated in equation 6.

RR

~
+ I

(6)

Effective Impedance = R F

F

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TPA711
70()..mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
component selection (continued)
As an example consider an input resistance of 10 kQ and a feedback resistor of 50 kQ. The BTL gain of the
amplifier would be -1 0 VN and the effective impedance at the inverting terminal would be 8.3 kQ, which is well
within the recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kQ, the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kQ. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.

~dB~====~~-----fc(lowpaSs)

1
(7)

For example, if RF is 100 kQ and CF is 5 pF, then fe is 318 kHz, which is well outside of the audio range.

input capacitor, C,
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, C, and R, form a high-pass filter with the comer frequency
determined in equation 8.

fc(highpaSS) =

2lt~ICI

(8)

The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where R, is 10 kQ and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
(9)

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TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION iNHiRiWATiON

component selection (continued)
In this example, CI is 0.40 /IF, SO one would likely choose a value in the range of 0.47 /IF to 1 /IF. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI' CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage at the inputtothe amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications, as the dc level there is held at Vool2, which is likely higher
than the source dc level. It is important to confirm the capacitor polarity in the application.
power supply decoupling, Cs
The TPA711 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also prevents
oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved
by using two capacitors of different types that target different types of noise on the power supply leads. For
higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESR)
ceramic capacitor, typically 0.1 /IF placed as close as possible to the device Voo lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capaCitor of 10 /IF or greater placed near the audio
power amplifier is recommended.
midrail bypass capaCitor, CB
The midrail bypass capacitor, CB, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CB determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR
THO + N. The capacitor is fed from a 250-kn source inside the amplifier. To keep the start-up pop as low as
pOSSible, the relationship shown in equation 10 should be maintained. This insures the input capaCitor is fully
charged before the bypass capaCitor is fuly charged and the amplifier starts up.
10
(C B x 250

kn)

s

1
(RF

+ RI)

CI

(10)

As an example, consider a circuit where CB is 2.2 /IF, CI is 0.47 /IF, RF is 50 kn, and RI is 10 k.Q. Inserting these
values into the equation 10 we get:
18.2 s 35.5
which satisfies the rule. Bypass capacitor, CB, values of 0.1 /IF to 2.2 /IF ceramic or tantalum low-ESR capacitors
are recommended for the best THO and noise performance.

single-ended operation
In SE mode (see Figure 51), the load is driven from the primary amplifier output (Vo+, terminal 5).
In SE mode the gain is set by the RF and RI resistors and is shown in equation 11. Since the inverting amplifier
is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included.
SE Gain = -

(~~)

(11 )

~TEXAS

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TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS23OB - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
component selection (continued)
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the following relationship:
10

1

<

..,;_1_

(C e X250kO)-(R F +R I)C I

(12)

RLCC

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier, thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 13.

fC(high)

(13)

Ie

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 I1F is chosen and loads vary from 4 0,
80, 320, and 47 kn. Table 2 summarizes the frequency response characteristics of each configuration.

Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
Cc

LOWEST FREQUENCY

ao

330ILF

60Hz

320

330ILF

15 Hz

47,0000

330 1LF

0.Q1 Hz

RL

As Table 2 indicates, an 8-0 load is adequate, earphone response is good, and drive into line level inputs (a
home stereo for example) is exceptional.

~TEXAS

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TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

SE/BTL operation
The ability of the TPA711 to easily switch between BTL and SE modes is one of its most important cost-saving
features. This feature eliminates the requirement for an additional earphone amplifier in applications where
internal speakers are driven in BTL mode but external earphone or speaker must be accommodated. Internal
to the TPA711 , two separate amplifiers drive VO+ and V0-. The SElBTL input (terminal 3) controls the operation
ofthe follower amplifier that drives Vo- (terminalS). When SE/BTL is held low, the amplifier is on and the TPA711
is in the BTL mode. When SElBTL is held high, the Vo- amplifier is in a high output impedance state, which
configures the TPA711 as an SE driver from VO+ (terminal 5). 100 is reduced by approximately one-half in SE
mode. Control of the SE/BTL input can be from a logic-level TTL source or, more typically, from a resistor divider
network as shown in Figure 52.

4

IN

VO+ 5

Cc

2 BYPASS

Vcr 8
7

GND

1 SHUTDOWN
3 SE/BTL
0.11J.F

T

100 IUl

VDD----~v-~------------------------------------------------~

100 IUl

Figure 52~ TPA711 Resistor Divider Network Circuit
Using a readily available 1/S-in. (3.5 mm) mono earphone jack, the control switch is closed when no plug is
inserted. When closed, the 1OO-knt1-kn divider pulls the SElBTL input low. When a plug is inserted, the 1-kn
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the Vo- amplifier is shut
down causing the BTL speaker to mute (virtually open-circuits the speaker). The Vo+ amplifier then drives
through the output capacitor (Cc) into the earphone jack.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capaCitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

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TPA711
7GO-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S230B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
5-V versus 3.3-V operation
The TPA711 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability.
The most important consideration is that of output power. Each amplifier in TPA711 can produce a maximum
voltage swing of Voo - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as
opposed to VO(PP) =4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into
an 8-0 load before distortion becomes significant.
Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power of operation from 5-V supplies for a given output-power level.

headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA711 data sheet, one can see that when the TPA711
is operating from a 5-V supply into a 8-0 speaker that 700 mW peaks are available. Converting watts to dB:
PdB = 10L09(:W) = 10Log
ref

(7~:W) = -1.5 dB

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
-1.5 dB -15 dB = -16.5 (15 dB headroom)
-1.5 dB -12 dB = -13.5 (12 dB headroom)
-1.5 dB - 9 dB = -10.5 (9 dB headroom)
-1.5 dB - 6 dB = -7.5 (6 dB headroom)
-1.5 dB - 3 dB = -4.5 (3 dB headroom)
Converting dB back into watts:
Pw = 10PdB j10 x P
ref
= 22 mW (15 dB headroom)
= 44 mW (12 dB headroom)
= 88 mW (9 dB headroom)
= 175 mW (6 dB headroom)
= 350 mW (3 dB headroom)

~.TEXAS

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TPA711
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS230B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION iNFORMATiON

headroom and thermal considerations (continued)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with 0 dB
of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings
for the system. Using the power dissipation curves for a 5-V, 8-0 system, the internal dissipation in the TPA711
and maximum ambient temperatures is shown in Table 3.
Table 3. TPA711 Power Rating, S-V,
PEAK OUTPUT
POWER
(mW)

AVERAGE OUTPUT
POWER

POWER
DISSIPATION
(mW)

a-o, BTL

DPACKAGE
(SOIC)

DGNPACKAGE
(MSOP)

MAXIMUM AMBIENT
TEMPERATURE

MAXIMUM AMBIENT
TEMPERATURE
110°C

700

700mW

675

34°C

700

350 mW (3 dB)

595

47"C

115°C

700

176mW(6dB)

475

68°C

122°C

700

88 mW (9 dB)

350

89°C

125°C

700

44 mW (12 dB)

225

111°C

125°C

Table 3 shows that the TPA711 can be used to its fuIl700-mW rating without any heat sinking in still air up to
110°C and 34°C for the DGN package (MSOP) and D pacakge (SOIC) respectively.

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3-204

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231 B - NOVEMBER 1998 - REVISED MARCH 2000

u OR j)GN PACKAGE

• Fully Specified for 3.3-V and S-V Operation

(TOP VIEW)

• Wide Power Supply Compatibility
2.5V-S.SV

=
=

Vcr

SHUTDOWN
BYPASS

=

• Output Power for RL 8 n
- 700 mW at Voo S V, BTL
- 250 mW at Voo 3.3 V, BTL

GND

Voo
Vo+

• Integrated Depop Circuitry
• Thermal and Short-Circuit Protection
• Surface-Mount Packaging
- SOIC
- PowerPADTM MSOP

description
The TPA721 is a bridge-tied load (BTL) audio power amplifier developed especially for low-voltage applications
where internal speakers are required. Operating with a 3.3-V supply, the TPA721 can deliver 2S0-mW of
continuous power into a BTL 8-n load at less than 0.6% THD+N throughout voice band frequencies. Although
this device is characterized out to 20 kHz, its operation was optimized for narrower band applications such as
wireless communications. The BTL configuration eliminates the need for external coupling capacitors on the
output in most applications, which is particularly important for small battery-powered equipment. This device
features a shutdown mode for power-sensitive applications with a supply current of 71JA during shutdown. The
TPA721 is available in an 8-pin sOle surface-mount package and the surface-mount PowerPAD MSOP, which
reduces board space by SO% and height by 40%.

Voo 6
RF

J.

Audio
Input

~C

RI

~

I

4

IN-

3

IN+

2

BYPASS

Vo0J2

r
, -,
,
,
,

,
,
,
,
,

CBr

Fro m System Control

4

~

'Y

SHUTOOWN

I
I

'-r

,

Bias
Control

-::-

~
Jyy

,

1

Cs

-.V

,
,

-

•

±

VOO

~

V

VO+ 5

J

1

1

Vcr

ajf
.........

700mW

7
GNO

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

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Copyright © 2000, Texas Instruments Incorporated

3-205

TPA721
70D-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231B- NOVEMBER 1998- REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
MSOpt

MSOP
SymbolIZatIon

TA

SMALL OUTLINEt
(D)

(OON)

-40°C to 85°C

TPA7210

TPA7210GN

ABC

t In the 0 package, the maximum output power is themally limited to 350 mW; 700 mW peaks
can be driven, as long as the RMS value is less than 350 mW.
:I: The 0 and DGN packages are available taped and reeled. To order a taped and reeled part, add
the suffIX R to the part number (e.g., TPA301 DR).

Terminal Functions
TERMINAL
NAME

NO.

110

DESCRIPTION

I

BYPASS is the tap to the voltage divider for intarnai mid-supply bias. This teminal should be connected
to a O.l-I1F 10 2.2-I1F capacitor when used as an audio amplifier.

BYPASS

2

GNO

7

IN-

4

I

IN+

3

I

IN+ is the noninverting input. IN + is typically tied 10 the BYPASS temina!.

SHUTDOWN

1

I

SHUTDOWN places the entire device in shutdown mode when held high (IDO < 711A).

VDO

6

VO+

5

0

VO+ is the positive BTL output.

Vo-

8

0

Vo- is the negative BTL output.

GND is the ground connection.
IN- is the inverting input. IN-is typically used as the audio input teminai.

VDD is the supply voltage leminal.

absolute maximum ratings over operating free-air temperature range {unless otherwise noted}§
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. ~5°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maximum ratings' may cause pemanent damage to the device. These are slress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions· is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE

=

=

PACKAGE

TAS25°C

DERATING FACTOR

D

725mW

5.8mWloC

TA 70°C
464mW

TA 85°C
377mW

DGN

2.14 w1I

17.1 mWrC

1.37W

1.11W

~ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package ApplICation Report

(literature number SLMAOO2), for more information on the PowerPAD package. The themal data was
measured on a PCB layout based on the infomatlon in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VDO
Operating free-air temperature, TA

-!I
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MIN

MAX

2.5

5.5

V

-40

85

°C

UNIT

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231 B - NOVEMBER 1998 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, Voo =3.3 V, fA =25u C (uniess otherwise
noted)
PARAMETER

TEST CONDITIONS

Voo
PSRR

output offset voltage (measured differentially)

See Note 1

Power supply rejection ratio

VOO=3.2Vt03.4 V

100

Supply current

BTL mode

IOO/SO)

Supply current. shutdown mode (see Figure 4)

MIN

TYP

MAX

UNIT

20

mV

1.25

2.5

rnA

7

50

ItA

dB

85

NOTE 1: At 3 V < VOO < 5 V the dc output voltage is approximately Vo0f2.

operating characteristics,

Voo = 3.3 V, TA = 25°C, RL = 8 n

PARAMETER
Po
THO+N
80M
Bl
ksVR
Vn

TEST CONDmONS

MIN

Output power. see Note 2

THO = 0.5%.

See Figure 9

Totel harmonic distortion plus noise

Po=250mW.

1= 200 Hz to 4 kHz. See Figure 7

TYP

MAX

250

UNIT
mW

0.55%

Maximum output power bandwidth

Gain =2.

THO = 2%.

Unity-gain bandwidth

Open Loop.

See Figure 15

See Figure 7

20

kHz

1.4

MHz

Supply ripple rejection ratio

1=1 kHz.

CB=lI1F•

See Figure 2

79

dB

Noise output voltege

Gain = 1.

CB = O.lI1F.

See Figure 19

17

IIV(rms)

NOTE 2: Output power is measured at the output terminals of the device at 1= 1 kHz.

electrical characteristics at specified free-air temperature,
noted)
PARAMETER

TEST CONDmONS

VOO
PSRR

Output offset voltage (measured differentially)

100

Supply current

IOO(SO)

Supply current. shutdown mode (see Figure 4)

Power supply rejection ratio

operating characteristics,

Voo = 5 V, TA = 25°C (unless otherwise
MIN

TYP

MAX

mV

1.25

2.5

rnA

50

100

ItA

dB

78

VOO=4.9VIo5.1 V

UNIT

20

Voo = 5 V, TA = 25°C, RL =8 n

PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

Output power

THO = 0.5%.

See Figure 13

700t

Total harmonic distortion plus noise

Po=250mW.

1= 200 Hz to 4 kHz. See Figure 11

0.5%

BOM

Maximum output power bandwidth

Gain =2.

THO =2%.

Bl

Unity-gain bandwidth

Open Loop.

See"Figure 16

kSVR

Supply ripple rejection ratio

1= 1 kHz.

CB=lI1F•

See Figura 2

80

dB

Vn

Noise output voltege

Gain = 1.

CB = O.lI1F•

See Figure 20

17

IIV(rms)

Po
THO+N

See Figure 11

mW

20

kHz

1.4

MHz

t The OGN package. properly mounted. can conduct 700 mW RMS power continuously. The 0 package can only conduct 350 mW RMS power
continuously wtih peaks to 700 mW.

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TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B- NOVEMBER 1998- REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

Voo 6

l

RF
Audio
Input

~C

RI

L

4

IN-

3

IN+

I

2

BYPASS

,r - - ,
,
,
,
,
,
,
,
,
'-~
,

,

CB -: =-

T

,
,
1

.V

~

I

SHUTOOWN

J-

I
I

r

VOO

i

Vo0J2

es

VO+ 5

RL=8

V(T" 8

- •

: V

BI8S1
Control

7
GNO

~

Figure 1. BTL Mode Test Circuit

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
ksVR

Supply ripple rejection ratio

vs Frequency

100

Supply current

vs Supply voltage

Po

Output power

THO+N

Total harmonic distortion plus noise

vs Supply voltage

5

vs Load resistance
vs Frequency

6
7,8,11,12

vs Output power

9,10,13,14

Open loop gain and phase

vs Frequency

15,16

Closed loop gain and phase

vs Frequency

17,18

Vn

Output noise voltage

vs Frequency

19,20

Po

Power dissipation

vs Output power

21,22

~TEXAS

INSTRUMENTS
3-208

2
3,4

POST OFACE BOX 655303 • OALLAS, TEXAS 75265

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231 B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY

SUPPLY CURRENT
vs
SUPPLY VOLTAGE

0
III

"
I

Q

Ic

~CD

1.&
RL=&f.I
CB=1/-lF
BTL

-10
-20

1.6
c(

-30

E
I

-40

'E
~
:::I

I -so
t
f -60

aa.

(J

1.2

:::I

----

~

-

~~

I/)

-70

'l1li:

:::I

'?a:
...i:i

aa.

./

1.4

~TL

-80

I

t...,...; ~

~OO=3.3V

r-..

C

E
0.&

VOO=5V

-90

-I ""11 I

-100
20

100

1k
f - Frequency - Hz

0.6
2.5

10k 20k

3

3.5

4

4.5

5

5.5

VDD - Supply Voltage - V

Figure 2

Figure 3
SUPPLY CURRENT
vs
SUPPLY VOLTAGE

90
SHUTDOWN

=High

&0

/
V

70
c(
:::I.

I

'E
~:::I

60
50

(J

~

a.
a.
:::I

I/)

I

40
30

C

E

20
10

o~
2.5

~
3

3.5

/

/

/

4

/

/

4.5

5

5.5

VOD - Supply Voltage - V

Figure 4

~TEXAS

INSTRUMENTS
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3-209

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

vs
SUPPLY VOLTAGE
THO+N1%
f=1kHz

BTL

~
I

I

J

400

I--_+-_~L--+-

I

,p

o~--~--~--~--~--~--~

2.5

3

3.5
4
4.5
5
VOO - Supply Voltage - V

5.5

FigureS
OUTPUT POWER

vs
LOAD RESISTANCE
800

700

~

600

I

500

f
0

I

,p

BTL

'\
\.VOO=5V

I

'S

THO+N=1%
1=1 kHz

\

" "'-

400

300 ~OO=3.3V

~

......

200

" ..............

100

o

r--.....

8

16

24

r--

32

40

---

48

56

RL - Load Resistance - 0

Figure 6

~TEXAS

3-210

INSTRUMENTS
POST OFFICE BOX 655303 • OALIJ\S, TEXAS 75265

64

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231 B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10
';I.

31

r-

'0

z

+
c

~

vs

FREQUENCY

FREQUENCY
10

r=

VOO=3.3V
~ PO=250mW
RL=8n
BTL

I

.~

IIII
~~~2OVN
1/

- .-!'"

~

as

0.1

j

Iz

"/

"

+

~

c
0

Po=~OmW

i:
0

~
-l!0

Ir

A

r!.

0

~ VOO=3.3V
~ RL=8n
_ AV=-2VN
_ BTL

I

1//

1

Jill

I

';I.

AV=-10VN

::c

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

~

/.'t'

~

AV =-2 VN

as

::c

~

0.1

PO=125mW

S

=

~

I

I

Z

~

Z

r-

0

+

Q

j!:

j!:

0.01

100

20

1k

10k

PO=250 mW
0.01

20k

100

20

FigureS

TOTAL HARMONIC DISTORTION PLUS NOISE

10

f=

1=

I

Iz

rr-

+

vs

OUTPUT POWER

OUTPUT POWER
10

VOO=3.3V
f=1 kHz
AV=-2VN
BTL

';I.
I

iz

1/

~

+
c

/

~

f=20kHz

::-,.......r.

~

F

~

I

j

f=10kHz

~

is

-l!0

::c

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

c

~
III

20k

f - Frequency - Hz

Figure 7

';I.

10k

1k

f - Frequency - Hz

--

0.1

~

RL=8n

.2

I

F;;;;;;:::.f=1kHz

c

.... -,...

0

I

S

I'I

~
I

Z

0.1

F

f=2OHz

VOO = 3.3 V
RL=8n
CB=1I1F
AV =-2 VN
BTL

~

+

Q

Q

::c
I0.01

j!:

o

0.05

0.1

0.15

0.2

0.25

0.3

0.35

0.4

0.01
0.01

Po - Output Power - W

I

I

I I II

0.1
Po - Output Power - W

Figure 9

Figure 10

-!II
TEXAS
INSTRUMENTS
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3-211

TPA721
70D-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231 B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
VB
FREQUENCY

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

10

~

VDD=5V
Po = 700 mW
RL=SO
BTL

I

Iz

+
c

Iz

1/'./

AV=-20~~"

0

~

.

-

+

Q

:z:
I-

Po=350mW -

I

-

Z

+

Q

:z:
I-

0.01
20

100

1k

10k

0.01

20k

20

.

rrr-

+

10

--

I

Iz

/

BTL

I

c
0

t:

I
I

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
OUTPUT POWER
~

~ VDD=5V
f=1 kHz
AV=-2VN

Iz

!"'"--.....

+

c

~
~

is

.S!

t--

RL=80~

:!

0.1

~

:z:

Q

I-

j!:

+

0.2

0.3

OA 0.5

0.6

0.7

0.8

0.9

1

f= 20 Hz

0.1

I

Z

0.01
0.01

r--- t -

VDD=5V
RL=80
CB=1I1F
AV =-2 VN
BTL

Po - Output Power - W

0.1
Po - Output Power - W

Figure 13

Figure 14

~TEXAS

3-212

f=20kHz

f= 10 kHz

I I j"j

j

I

Z

1-1"-

I-

'=1 kHz

~

j
{!!.

0.01
0.1

-r-

~0

c
0

20k

Figure 12

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
r-

10k

f - Frequency - Hz

Figure 11

10

1k

100

f - Frequency - Hz

~
I

~

Po = 700 mW

0.1

./

r. . .

A

I "
j

A~2VN

j
I

~

~

A

~~

Z

~

~

I

AV=-10Y.,N

j'-.

0.1

Po=50mW.;

BTL

c

~
:z:

r-

+

/

t:0
r-

~ VDD=5V
~ RL=80
r- AV=-2VN

I

-

.S!
c

10

~

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY

80
70

60
III

"I

~

180"
VOO=3.3V
RL=Open
BTL

"-

30

Gain

Go

~!.

0

i"""--~

~

40

140°

r- 100°

Phase

50

r

20

~

10

"-

0

-600
~

-1000

-10

r- -140"

-20

104

_180°

f - Frequency - kHz

Figure 15
OPEN-LOOP GAIN AND PHASE

vs
FREQUENCY

80
70

60

180°
VOO=5V
RL=Open
BTL

"Phase

III

"cI

~

8

~
!.

0

50

.......

~

40

140°

1000

,

60°

20°

30

Gal~

20
10

"-

0

J

_2001L

,

_100°

-10
_140°

-20
-30
1

101
f - Frequency - kHz

103

104

-180"

Figure 16

~TEXAS

INSTRUMENTS
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3-213

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231B- NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY

V

0.75
0.5
III
"U

I

iCJ
Co

0

i
(j

Phase .........

/

/

0.25

"\

\

I

0
-0.25

{

-0.5

/'

\.

Gain

\

\
\
\

-0.75
-1
-1.25

VOO=3.3V
RL=8f.l
Po = 250 mW
BTL
I

-1.5
-1.75

-2

102

101

~.

,
103

104

1\
\

105

f - Frequency - Hz

Figure 17
CLOSED-LOOP GAIN AND PHASE

vs
FREQUENCY
180"
",....- Phase -

0.75
0.5
III
"U
I

c

0.25
0

~

-0.25

0

-0.5

I

Co·

.9

..
0

i

L
/

{

"\ \

\

/'

Gain

""\

\

-0.75

\

-1
-1.25
-1.5
-1.75

-2
101

\
\

VOO=5V
RL=8f.l
Po=700mW
BTL
104
f - Frequency - Hz

Figure 18

~TEXAS

3-214

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231B - NOVEMBER 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

OUTPUT NOISE VOLTAGE

vs

vs

FREQUENCY

100

~

FREQUENCY

100

: VOO = 3,3 V
BW = 22 Hz to 22 kHz
RL=Snor32n
AV=-l VN

tI

~

CD
III

t
.

VOBTL

:!!!
z

RL = Snor32 n

r- AV=-l VN

~

I

CD
CI

'0

1= VOO'; S'V

1= BW = 22 Hz to 22 kHz

VOBTL

~

Vo+

til

10

'0

z

'5

Vo+

10

'5

~

~

0

0

I

I

C

-:f

>

1

20

lk

100

1
20

10k 20k

lk

100

f - Frequency - Hz

10k

Figure 19

Figure 20

POWER DISSIPATION

POWER DISSIPATION

vs

vs

OUTPUT POWER

OUTPUT POWER

~O~------~---------r--------'

800

BTLM~e

300 r-----j;;;;;;;;;o;;;;;;;..!'1

700

VOO=5V

BTL Mode
Voo=3,3V

260~--~---+---------r~~~--i

200~4------+---------r--------i

3=

600

i

500

E
I
c

.!!!

c

~

300

I

200

/

400

600

o

./

~

~

I

1

L

100

-

I

RL=sn -

V

400

a.~

~

200

20k

f - Frequency - Hz

o

Po - Output Power - mW

Figure 21

~2n

~ r-....
200

400
SOO
800
Po - Output Power - mW

1000

Figure 22

~TEXAS

INSTRUMENTS
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3-215

TPA721
700·mW MONO LOW·VOLTAGE AUDIO POWER AMPLIFIER
SL0S231B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
bridged-tied load
Figure 23 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA721 BTL amplifier consists
of two linear amplifiers driving both ends of the load. There are several potential benefits to this differential drive
configuration but initially consider power to the load. The differential drive to the speaker means that as one side
is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage swing on the
load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where voltage is
squared, yields 4x the output power from the same supply rail and load impedance (see equation 1).
VO(PP)
V(rms) =
Power

2/2
V(rms)

(1)

2

=RL
Voo

J' ;

RL

J'!
'V;

VO(PP)

2x vO(PP)

-VO(PP)

Figure 23. Bridge-Tied Load Configuration

In a typical portable handheld equipment sound channel operating at 3.3 V, bridging raises the power into an

a-n speaker from a singled-ended (SE, ground reference) limit of 62.5 mW to 250 mW. In sound power that is
a 6-dB improvement - which is loudness that can be heard. In addition to increased power, there are frequency
response concerns. Consider the single-supply SE configuration shown in Figure 24. A coupling capacitor is
required to block the dc offset voltage from reaching the load. These capacitors can be quite large
(approximately 33 J.1F to 1000 J.1F) so they tend to be expensive, heavy, occupy valuable PCB area, and have
the additional drawback of limiting low-frequency performance of the system. This frequency limiting effect is
dl'e to the high pass filter network created with the speaker impedance and the coupling capacitance and is
calculated with equation 2.
(2)

f(comer)

-!111ExAs

3-216

INSTRUMENTS
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TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
bridged-tied load (continued)
For example, a 68-J.lF capacitor with an 8-il speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

VOO

~dB~-----J~=====

Figure 24. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of a SE
configuration. Intemal dissipation versus output power is discussed further in the thermal considerations
section.

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from Voo.
The internal voltage drop multiplied by the RMS value of the supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out being equal to the ratio of power from the power supply
to the power delivered to the load. To accurately calculate the RMS values of power in the load and in the
amplifier, the current and voltage waveform shapes must first be understood (see Figure 25).
100

,/

V(LRMS)

---fVVVVffl'l-

IOO(FiMS)

Figure 25. Voltage and Current Waveforms for BTL Amplifiers

~ThxAs

INSTRUMENTS
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TPA721
7DO-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
Sl0S231B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
Efficiency =

P"'P =-

(3)

SUP

Where:

= VOO loorms

2Vp
= 1t RL
1t

p R
( -'=--.b

)1/2

2

1tV

(4)

Efficiency of a BTL Configuration = 2V P

DO
Table 1 employs equation 4 to calculate efficiencies for three different output power levels. The efficiency of the
amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resultIng in a·
nearly flat internal power dissipation over the normal operating range. The internal dissipation at full output
power is less than in the half power range. Calculating the efficiency for a specific system is the key to proper
power supply design.
Table 1. Efficiency vs Output Power In 3.3-V 8-0 BTL Systems
OUTPUT POWER

EFFICIENCY

(W)

(%)

0.125

33.6

PEAK-to-PEAK
VOLTAGE
(V)

INTERNAL
DISSIPATION

1.41

0.26

47.6
2.00
0.25
2.45t
58.3
0.375
t High-peak voltage values cause the THO to increase.

(W)

0.29
0.28

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. In equation 4, Voo is in the denominator. This indicates
that as Voo goes down, efficiency goes up.

~1EXAS

3-218

INSTRUMENTS
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TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
application schematic
Figure 26 is a schematic diagram of a typical handheld audio application circuit, configured for a gain of
-10VN.

RF
Audio
Input

RI

~c

L

10kn

VDD 6

..1\,

~

50kn

4

IN-

3

IN+

2

BYPASS

I

I

,r
,

,
,
,
,
,
,

T

,
,

-=-

,
1

SHUTDOWN

--

,
,

CB -:::::2.211F

From System Control

-l. Cs

VD0f2

I
I

-.V

~

'-~

:V
r -.

Bias
Control

T

VDD

11JF

Vo+ 5

J

1

I

Vo-

e=rr-...

700mW

7

GND

~

Figure 26. TPA721 Application Circuit
The following sections discuss the selection of the components used in Figure 26.

component selection
gain setting resistors, RF and RI
The gain for each audio input of the TPA721 is set by resistors AF and AI according to equation 5 for BTL mode.
BTL Gain = -

2(~~)

(5)

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA721 is a MOS amplifier, the input impedance is very high;
consequently input leakage currents are not generally a concern, although noise in the circuit increases as the
value of AF increases. In addition, a certain range of AF values is required for proper startup operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kn and 20 kn. The effective impedance is calculated in equation 6.
Effective Impedance

=

A

AA
~~
F

(6)

I

~1ExAs

INSTRUMENTS
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3-219

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231 B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
gain setting resistors, RF and R, (continued)
As an example consider an input resistance of 10 ill and a feedback resistor of 50 kn. The BTL gain of the
amplifier would be -1 0 VN and the effective impedance at the inverting terminal would be 8.3 kn, which is well
within the recommended range.
For high performance applications, metal film resistors are recommended because they tend to have lower
noise levels than carbon resistors. For values of RF above 50 kO, the amplifier tends to become unstable due
to a pole formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kn. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.

~dBF=====~~-----(7)

f co(lowpass)

Ie

For example, if RF is 100 ill and CF is 5 pF, then fco is 318 kHz, which is well outside of the audio range.

input capacitor, C,
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, C, and R, form a high-pass filter with the corner frequency
determined in equation 8.

fcO(highpass) =

23t~ICI

(8)

The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where R, is 10 ill and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
(9)

~TEXAS

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TPA721
700·mW MONO LOW·VOLTAGE AUDIO POWER AMPLIFIER
SLOS231 B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, CI (continued)
In this example, CI is 0.40 I1F, so one would likely choose a value in the range of 0.47 I1F to 1 I1F. A further
consideration for this capacitor is the leakage path from the input source through the input network (AI, CI) and
the feedback resistor (AF) to the load. This leakage current creates a dc offset voltage at the input to the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. It is important to confirm the capacitor polarity in the application.
power supply decoupling, Cs
The TPA721 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling to
ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also prevents
oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is achieved
by using two capacitors of different types that target different types of noise on the power supply leads. For
higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance (ESA)
ceramic capacitor, typically 0.1 I1F placed as close as possible to the device Voo lead, works best. For filtering
lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 I1F or greater placed near the audio
power amplifier is recommended.
midrail bypass capacitor, CB
The mid rail bypass capacitor, Ce, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, Ce determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit intemal to the amplifier, which appears as degraded PSAA and
THD + N. The capacitor is fed from a 250-kn source inside the amplifier. To keep the start-up pop as low as
pOSSible, the relationship shown in equation 10 should be maintained. This insures the input capacitor is fully
charged before the bypass capacitor is fully charged and the amplifier starts up.
10
(C B

x 250

<

1

kn) - (RF + RI) CI

(10)

As an example, consider a circuit where Cs is 2.2I1F, CI is 0.47I1F, RF is 50 kn, and RI is 10 kn. Inserting these
values into the equation 10 we get:
18.2 s 35.5
which satisfies the rule. Bypass capacitor, Ce, values of 0.1 I1F to 2.2 I1F ceramic or tantalum low-ESR
capacitors are recommended for the best THD and noise performance.

using low-ESR capaCitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capaCitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capaCitor in the circuit. The lower the equivalent value of this
reSistance, the more the real capaCitor behaves like an ideal capaCitor.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-221

TPA721
70D-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SLOS231B- NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
5-V versus 3.3-V operation
The TPA721 operates over a supply range of 2.5 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation with respect to supply bypassing, gain setting, or stability.
The most important consideration is that of output power. Each amplifier in TPA721 can produce a maximum
voltage swing of VDD - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as
opposed to VO(PP) 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into
an 8-0 load before distortion becomes significant.

=

Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power than operation from 5-V supplies for a given output-power level.

headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA721 data sheet, one can see that when the TPA721
is operating from a 5-V supply into a 8-0 speaker that 700 mW peaks are available. Converting watts to dB:
PdB

=

10LogPW

= 10Log 700

mW

= -1.5

dB

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
-1.5 dB - 15 dB = -16.5 (15 dB headroom)
-1.5 dB -12 dB =-13.5 (12 dB headroom)
-1.5 dB - 9 dB = -10.5 (9 dB headroom)
-1.5 dB - 6 dB = -7.5 (6 dB headroom)
-1.5 dB - 3 dB = -4.5 (3 dB headroom)
Converting dB back into watts:
Pw

=
=
=

10 PdB /10
22 mW (15 dB headroom)
44 mW (12 dB headroom)

= 88

mW (9 dB headroom)

175 mW (6 dB headroom)

= 350 mW (3 dB headroom)

~TEXAS

3-222

INSTRUMENTS
POST OFFICE BOX 856303 • DAllAS, TEXAS 75265

TPA721
700-mW MONO LOW-VOLTAGE AUDIO POWER AMPLIFIER
SL0S231 B - NOVEMBER 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
headroom and thermal considerations (continued)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 700 mW of continuous power output with 0 dB
of headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings
for the system. Using the power dissipation curves for a 5-V, 8-n system, the internal dissipation in the TPA721
and maximum ambient temperatures is shown in Table 2.
Table 2. TPA721 Power Rating, 5-V, 8-n., BTL
PEAK OUTPUT
POWER

(mW)

AVERAGE OUTPUT
POWER

POWER
DISSIPATION

(mW)

DPACKAGE
(SOIC)

DGNPACKAGE
(MSOP)

MAXIMUM AMBIENT
TEMPERATURE
(OCFM)

MAXIMUM AMBIENT
TEMPERATURE
(OCFM)

700

700mW

675

34°C

110°C

700

350 mW (3 dB)

595

47°C

115°C

700

176 mW {6 dB)

475

68°C

122°C

700

88 mW (9 dB)

350

89°C

125°C

700

44 mW(12 dB)

225

111°C

125°C

Table 2 shows that the TPA721 can be used to its full 700-mW rating without any heat sinking in still air up to
110°C and 34°C for the DGN package (MSOP) and D pacakge (SOIC) respectively.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-223

3-224

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOS164A - SEPTEMBER 1996 - REVISED MARCH 2000

• 1-W BTL Output (S Y, 0.2 % THD+N)

DPACKAGE
(TOP VIEW)

• 3.3-Y and S-Y Operation
• No Output Coupling Capacitors Required

GND
SHUTDOWN
HP-SENSE
GND
BYPASS
HP-IN1
HP-IN2
GND

=

• Shutdown Control (100 0.6 IlA)
• Headphone Interface Logic
• Uncompensated Gains of 2 to 20 (BTL
Mode)
• Surface-Mount Packaging
• Thermal and Short-Circuit Protection

GND

V02
IN+
IN-

Voo
GAIN
Vo1
GND

• High Power Supply Rejection
(S6-dB at 1 kHz)
• LM4860 Drop-In Compatible

description
The TPA4860 is a bridge-tied load (BTL) audio power amplifier capable of delivering 1 W of continuous average
power into an 8-0 load at 0.4 % THD+N from a S-Y power supply in voiceband frequencies (f < 5 kHz). A BTL
configuration eliminates the need for external coupling capacitors on the output in most applications. Gain is
externally configured by means of two resistors and does not require compensation for settings of 2 to 20.
Features of this amplifier are a shutdown function for power-sensitive applications as well as headphone
interface logic that mutes the output when the speaker drive is not required. Internal thermal and short-circuit
protection increases device reliability. It also includes headphone interface logic circuitry to facilitate headphone
applications. The amplifier is available in a 16-pin sOle surface-mount package that reduces board space and
facilitates automated assembly.

typical application circuit
VOD 12
r-~~r-~~~r-----------------~--___

RF

Audio
Input

Vo0J2

-=-

~~
-=-

11

GAIN

13

IN-

14

IN+

VDD

-:rCa

1W
CBr

-=-

VOD

Ne

r+
-=-

5

BYPASS

6
7

HP-IN1

3

HP-SENSE

2

SHUTDOWN

RpU

I
I

Headphone
Plug

-=-

HP-IN2

1,4,8,9,16

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

~TEXAS

copyright © 2000, Texas Instruments IrlCOfPOlBted

INSTRUMENTS
POST OFFICE BOX 855303 • DALLAS, TEXAS 75265

3-225

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A- SEPTEMBER 1996- REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICE
TA

SMALL OUTLINE
(D)

-4O"C to 85°C

TPA48600

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (See Dissipation Rating Table)
Operating free-air temperature range, TA ............................................ -40°C to 85°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t Stresses beyond those listed under "absolute maximum ratings" may cause pennanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maxlmum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
DERATING FACTOR

PACKAGE

o

10mW/oC

1250mW

800mW

650mW

recommended operating conditions
Supply voltage, VDO
Common-mode input voltage, VIC

II VOO = 3.3 V
VOO=5V

Operating free-air temperature. TA

~TEXAS

INSTRUMENTS
3-226

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

MIN

MAX

2.7

5.5

V

1.25

2.7

V

1.25

4.5

V

-40

85

°C

UNIT

TPA4860
1-W MONO AUDIO POWER AMPLIFIER
SLOSI64A - SEPTEMBER 1996 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature range, VDD
noted)
PARAMETER
VOO

= 3.3 V (unless otherwise

TEST CONDmONS

Output offset voltage (measured differentially)

See Note 1

Supply ripple rejection ratio

VOO = 3.2 V to 3.4 V

TPA4860
MIN

TYP

MAX

5

20

UNIT
mV

75

dB

2.5

rnA

100

Quiescent current

IOO(M)

Quiescent current, mute mode

750

tJA

IOO(SO)

Quiescent current, shutdown mode

0.6

tJA

VIH

High-level input voltage (HP-IN)

1.7

V

VIL

Low-level input voltage (HP-IN)

1.7

V

VOH

High-level output voltage (HP-SENSE)

10 = 100 tJA

VOL

Low-level output voltage (HP-SENSE)

10=-100tJA

2.5

V

2.8
0.2

0.8

V

NOTE 1: At 3 V < VOO < 5 V the dc output voltage is approximately Vool2.

operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 8 n
PARAMETER

Po

TEST CONDmoNS

Output power, see Note 2

THO = 0.2%,
AV=2

1= 1 kHz,

THO = 2%,
AV=2

1= 1 kHz,
THO=2%

TPA4860
MIN

TYP

MAX

UNIT

350

mW

500

mW

BaM

Maximum output power bandwidth

Gain = 10,

20

kHz

Bl

Unity-gain bandwidth

Open Loop

1.5

MHz

I BTL

1 = 1 kHz

56

dB

ISE

1= 1 kHz

30

dB

Gain =2

20

ltV

Supply ripple rejection ratio
Vn

Noise output voltage, see Note 3

NOTES: 2. Output power is measured at the output terminals 01 the device.
3. Noise voltage Is measured In a bandwidth of 20 Hz to 20 kHz.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-227

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A - SEPTEMBER 1996 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature range, Voo
noted)
,
PARAMETER
Voo

TEST CONDITIONS

Output offset voltage

See Note 1

Supply ripple rejection ratio

Voo =4.9Vfo 5.1 V

=5

V (unless otherwise
TPA4860

MIN

TYP

MAX

5

20

UNIT
mV

70

dB

100

Supply current

3.5

mA

IOO~Ml
IOOCSO)
V,H

Supply current, mute

750

I1A

Supply current, shutdown

0.6

IlA

High-level input voltage (HP-IN)

2.5

V

V,L

Low-level input voltage (HP-IN)

2.5

V

VOH

High-level output voltage (HP-SENSE)

10= 500 IlA

VOL

Low-level output voltage (HP-SENSE)

10 =-500 IlA

2.5

2.8
0.2

V
0.8

V

NOTE 1: At 3 V < VOO < 5 V the de output voltage is approximately voot2.

operating characteristic, Voo = 5 V, TA = 25°C, RL = 8.a
PARAMETER

TEST CONDITIONS
THO = 0.2%,

Po

1= 1 kHz,

kJ=2

Output power, see Note 2

THO = 2%,

1=1 kHz,

kJ=2
BOM
B1

MAX

UNIT

1000

mW

1100

mW

Maximum output power bandwidth

Gain = 10,

20

kHz

Open Loop

1.5

MHz

I BTL

f= 1 kHz

56

dB

ISE

f= 1 kHz

30

dB

Gain =2

20

I1V

Noise output voltage, see Note 3

NOTES: 2. OUtput power is measured at the output terminals of the device.
3. Noise voltage is measured in a bandwidth of 20 Hz to 20 kHz.

~1ExAs

3-228

1YP

Unity-gain bandwidth
Supply ripple rejection ratio

Vn

TPA4860
MIN

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

THO=2%

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A- SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE

Voo

Output offset voltage

Distribution

1,2

100

Supply current distribution

vs Free-air temperature

3,4

THD+N

vs Frequency

5,6,7,8,9,
10,11,15,
16,17,18

vs Output power

12,13,14,
19,20,21

Total harmonic distortion plus noise

100

Supply current

vs Supply voltage

Vn

Output nOise voltage

vs Frequency

Maximum package power dissipation

vs Free-air temperature

Power dissipation

vs Output power

Maximum output power
Output power

vs Free-air temperature

22
23,24
25
26,27
28

vs Load Resistance

29

vs Supply Voltage

30

Open loop frequency response

vs Frequency

31

Supply ripple rejection ratio

vs Frequency

32,33

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-229

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
DISTRIBUTION OF TPA4860
OUTPUT OFFSET VOLTAGE

DISTRIBUTION OF TPA4860
OUTPUT OFFSET VOLTAGE

25

20

20 I---+--+---If---t--

f

I.

I

15

E

CC

'0

j

15

1---+--+--+-

'0

j

10

E
::s

Z

5

10 1---+--+--+-

5

-2 -1

2

0

3

4

5

6

7

-2 -1

Voo - Output Offset Voltage - mV

0

2

3

Figure 1

=

Vee 3.3 V

4

I

3

3.5
3

'\

'\

2.5

(.)

~

a.
a.

::s
II)
I

/'

1

Jf

2.5

I

/

'E
~
::s

2

.\/'

\PIC~(

(.)

Typical

~

a.
a.

2

::s

1.5

/'

II)

1.5

I
Q
Q

Q
Q

0.5

0.5

0

0

-20

85

TA - Free-Air Temperature -

·c

Figure 3

-20

25

INSTRUMENTS
POST OFFICE BOX 656303 • DALLAS, TEXAS 75265

85

TA - Free-Air Temperature _·c

Figure 4

~TEXAS

3-230

7

3.5
VCC=5V

I

6

SUPPLY CURRENT DISTRIBUTION
vs
FREE-AIR TEMPERATURE

4.5

E

5

Figure 2

SUPPLY CURRENT DISTRIBUTION
vs
FREE-AIR TEMPERATURE

CC

4

Voo - Output Offset Voltaga - mV

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

.,.

.,.

VOO=5V
PO=1W
AV =-10 VN
r- RL=80

f=

f=
r-

I!!I
ii:

I

~

CB =0.1I1F

.!!

r-..
0.1

~

!'h....1

ill"

i
~

CB=1I1F

CB=0.1I1F

i3

Is

..... 10-

~

I lli'
CB =111F

0.1

{!!.

I

I

~
j!:

10

I

VOO=5V
~ PO=1W
f- AV=-2VN
r- RL=80

ii:

J

FREQUENCY

f:

I!!I

s~

vs

FREQUENCY
10

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

Z

~

j!:

0.01
20

10 k 20 k

1k

100

0.01
20

100

f - Frequency - Hz

FigureS

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

.,.

vs

FREQUENCY

FREQUENCY

10

==
=

I!!I

VOO=5V
PO=1W
AV=-2OVN _
RL=80
,_

CB=0.1I1F

ii:
c

f"

~

==

=

VOO=5V
PO=0.5W AV=-2VN
RL=80
,-

:
·0
z

!!I

u

CB=1I1F

·1

0.1

:J:

~B=0.1I1F

1000'

0.1

s

~

~

I
Z

'"

I

CB=1I1F

Z

~

j!:

10

I

I

.!!

Js

.,.
ii:

i

I-

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
I

10 k 20 k

1k
f - Frequency - Hz

~

j!:

0.01
20

100

1k

10k 20 k

0.01

20

f - Frequency - Hz

III I IIII
100

1k

10 k 20 k

f - Frequency - Hz

Figure 7

Figure 8

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-231

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOS164A- SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

~

vs

FREQUENCY

FREQUENCY

10

I

r--

i

i

"-

VOO=5V
PO=0.5W
1-r-~'Ioo.HH-~III--+---I--+++H1ff- AV = -20 VN _
1--I-+4f+,,-=CB =0.1 I1F
I- RL = 8 0
_

VOO=5V
PO=O.5W
AV=-10VN _
RL=80

J

_

==
10.~~~
=

==
=

I

Ij

TOTAL HARMONIC DISTORTION PLUS NOISE

va

-

CB=0.1I1F

"

......
~

t\
,./

0.1

"'"

CB=1I1F

I

~

i!=

0.01
20

100

1k

100

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

10

I

J

j

OUTPUT POWER
~
I

~

'I"--l 1""

_

CB=0.1I1F

'gB=1J
0.1

!

~
I

~

Z

0

I

J

!

RL=320
Po=60mW

I

:1:
....

f=2OHz

g

-,...
~ EE

VOO=5V
AV=-2VN
RL=80

it

PO=250mW

0.1

10

I!

II RL~80

I

I

vs

FREQUENCY

F
VOD=5V
r- Av=-10VN

I

TOTAL HARMONIC DISTORTION PLUS NOISE

va

~ Single Ended
II
I

I

0.01
20

i!=

100

1k

10k 20k

0.01
0.02

f - Frequency - Hz

0.1
Po - Output Power - W

Figure 12

Figure 11

~TEXAS

INSTRUMENTS

3-232

10k 20k

Figure 10

Figure 9

~

1k
f - Frequency - Hz

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

2

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOS164A - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

~
I

10

vs

OUTPUT POWER

OUTPUT POWER
~

= VOO=5V

=

:

-

..
f

g

c

~

Q
.2
c

!01

0.1

X

CB=0.1IlF

0.1

!

~

~

I

I

Z

Z

+
Q

c!
0.01
0.02

0.1

2

...X

0.01
0.02

Po - Output Power - W

Figure 13

Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

FREQUENCY

FREQUENCY
~

VOO=3.3V
Po = 350 mW
RL=8C
AV=-2VN

I
~c

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

I

2

0.1

Po - Output Power - W

10
VOO = 3.3 V
PO=350mW
RL=8C
AV=-10VN

I

I
~c
o

~

'E

i

.2

~B=0.1 J1F

X

1111"

I '"
f=
0.1

!

~-

I

~

.. -

-CB=1J1F

.S!

CB = 0.1 IlF
'low.

:!

III"

~

"

......

~

!

~

I

0.1

.......

......

~

~

=
- r- CB=1 IlF

I

~

j!:

II

~0

CB=0.1IlF

!

~

I'

r- ...

~0

i

j!:

:: VOO=5V
AV= 2VN
_ RL=8C
_ f= 20 kHz

=

Iz

ii:

X~OI

10

I

AV= 2VN
_ RL=8C
_ f=1kHz

~
!I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

~
0.01
20

j!:

100

1k

10k 20k

0.01
20

f - Frequency - Hz

100

1k

10 k 20k

f - Frequency - Hz

Figure 15

Figure 16

:II
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-233

_.-1
TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A - SEPTEMBER 1996 - REVISEO MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
VB
FREQUENCY
.,.

.,.

10
VOO=3.3V ::
PO=350mW RL=SO
AV=-20VN _

I

=

j
l

~ CB=0.1J,1F

I
J
_

0.1

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
FREQUENCY

1==

'"

~
!I

ii:

I
j

_

CB=1 J,IF

~

l

I

J

~
=

~

0.01
20

i!:
100

10k 20k

1k

10

1k

Figure 17

Figure 18

.,.

10k 20k

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
OUTPUT POWER
10

I

VOO=3.3V
r;:AV=-2VN
I- RL=SO
f=20Hz

::=

100

f - Frequency - Hz

1=

~

0.01
20

f - Frequency - Hz

.;0

F Av=-2VN
Voo=3.3V
I-

~

Z

RL=SO

I- f=1 kHz

II)

::s

ii:

111111

c

0

'E0

CB=0.1 J,IF

~

~

.2

c
0

E
01

0.1 ~ ~ CB=1.0J,IF

:c

-

0.1

~

!

~

--

CB=0.1I1F

I
Z

I

~

i!:

~l320

I

o

:c

0.1

XJ....

~

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
OUTPUT POWER

I

,

RL=SO
PO=250mW
Y'

Po=60mW

Z

I

I

........

.~

r--"':P'"
I'

~

.,.

VOO = 3.3 V
AV=-10VN
Single Ended

3l

!

F-

10

I

0.01
0.02

~
:c

I-

0.1
Po - Output Power - W

2

0.01
0.02

Figure 19

0.1
Po - Output Power - W

Figure 20

~TEXAS

INSTRUMENTS

POST OFFIC~X 655303 • DALlAS. TEXAS 75265

3-234

\

2

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

'#

SUPPLY CURRENT

vs

vs

OUTPUT POWER

SUPPLY VOLTAGE

10

5.---.---~----r----r---'----'

I

I

r-

Ic

I

1==1= Ce=0.1;F-

f

41----1--

V

i

1
::c

0.1

S

~ VDD=3.3V

~

AV=-2VN
RL=Sn
c- f:20kHz

I-

I

~

i!:

I
0.01
20m

I

I I III

0.1
Po - Output Power - W

O~--~--~----~--~--~--~

2

2.5

3

3.5
4
4.5
VDD - Supply Voltage - V

OUTPUT NOISE VOLTAGE

OUTPUT NOISE VOLTAGE

vs

vs

FREQUENCY

FREQUENCY
103

103

VCC=3.3V

VCC=5V

>:::I.

>

:::I.

I

I

!

t

102

102

~

~

Iz

V01 +V02 - I- F-V02

-

-" 11 I ==

'S

t

5.5

Figure 22

Figure 21

III
CI

5

101

0

V01

I

>c

-

i

r

J..

V02

~

101

0

V01

I

::f
1

1

20

V01 +V02

Iz

100

1k
f - Frequency - Hz

10k 20k

20

100

1k

10k 20 k

f - Frequency - Hz

Figure 23

Figure 24

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-235

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
MAXIMUM PACKAGE POWER DISSIPATION

POWER DISSIPATION

vs

vs

FREE-AIR TEMPERATURE

OUTPUT POWER

1.5

1.5
VOO=5V

;::
I

1.25

c
0

~

Iis

!

0.75

D.

~

II
aJ

.=
~

0.5

I

0.25

E
:I
E

I

I
I
I

1',

~

1',

~

\

0

25

50

75

100

V

0.5

'\

o
-25

/

;::

125

150

o

175

i

V

RL=40

- r--.

RL=80

V

~
o

I

~

r-RL=160

I I

0.25

0.5

TA - Free-Air Temperature - °C

0.75

1.25

Po - Output Power -

1.5

1.75

w

Figure 26

Figure 25
POWER DISSIPATION

MAXIMUM OUTPUT POWER

vs

vs

OUTPUT POWER

FREE-AIR TEMPERATURE

160
VOO=3.3V

140

;::

~

100

\\

.
:;:

80

I

c

---

RL=40

0

I
I

~.

I!!

I

0.5

~

is

0.25

o

ro

l
E
re.

RL=80

\

120

oU

0.75

~

~'.

60

1---

'.

I

1"RL=160

0.5

Po - Output Power -

0.75

w

o
o

0.25

RL=80

0.5

0.75

RL=40
I

1.25

Po - MaxImum Output Power - W

Figure 28

Figure 27

~1ExAs

INSTRUMENTS
3-236

--- ---

I'" ~r---

40
20

-I

0.25

,...--

\

IL.

.s-

~
RL=160

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

1.50

TPA4860
1·W MONO AUDIO POWER AMPLIFIER
SLOSl64A - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

vs

LOAD RESISTANCE

SUPPLY VOLTAGE

1.4

!

\.

\

a.
I

,p

;::

0.4

I

1\

0

I\.

"- ~CC=5V

'~ ~

0.2

4

.....

'5

1.251---+---+--+__---l----j~_,I

!
!

0.75

1----+----+~"--+_:'O"""--+----f---::;;;ooI

0.25

~=---::::b.......~--+__--+---j----l

a.
I

........

--

r- r- r-

,p
I--

Vcc = 3.3 V
(
I
I

o
8

12

AV=-2VN
f= 1 kHz
CB=0.1I-tF
THO+nS1%

1.75

1\

I

0.6

f=1 kHz
_
CB=0.1I-tF
THO+nS1%

\

;::
0.8

2.---~--~----.----r---'----'

A~=-4vNI

1.2

'5
ICL
'!i

OUTPUT POWER

vs

16 20 24 28 32 36
Load Resistance -!l

40 44 48

4.5
3.5
4
Supply Voltage - V

3

Figure 29

5

5.5

Figure 30
SUPPLY RIPPLE REJECTION RATIO

vs
FREQUENCY

OPEN LOOP FREQUENCY RESPONSE
100

80

III

"'"11.

i'\

60

iii 40
CI

II:

-30

I

c

L III

I

0

Phase

Gain

20

0°

-450

r,~

I

CI

"

VOO=5V
-10 I- RL=8!l
Bridge Tied
-20 I- Load

III

i
~

"cI

o

45°

Voo=5V
RL=8!l
CB=0.1I-tF

~

\

31
-90° .!
a.

I-

-135°

1\
0

'is

f
iii:

a

-40
-60

I

CB=0.1I-tF

I.

-70

I I

'-~B~1~~

-

..;

ICL
:::I

II)

~

-180°

1M

-225°
10M

-80
-90

-20
10

100

1k
10 k 100 k
f - Frequency - Hz

-100
100

Figure 31

1k
f - Frequency - Hz

10 k 20 k

Figure 32

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-237

TPA4860
1-W MONO AUDIO POWER AMPLIFIER
SLOS164A - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY

o
CD

"I

ig
i"i'

-10

-20

-40

a:

-so

ia:

...{JO

J

IIII

IC~~~.;II1F
....

[".. ......

f""... .......

VDD=5V
RL=8n

r--...... ........

.... :-.

-

Single Ended

r-:-.

~ ........

l7

CS=1I1F

-70
...{JO

-90

-100
100

1k

10k 20k

f - Frequency - Hz

Figure 33

APPLICATION INFORMATION
bridged-tied load versus single-ended mode
Figure 34 shows a linear audio power amplifier (APA) in a bridge tied load (BTL) configuration. A BTL amplifier
actually consists of two linear amplifiers driving both ends of the load. There are several potential benefits to
this differential drive configuration but initially let us consider power to the load. The differential drive to the
speaker means that as one side is slewing up the other side is slewing down and vice versa. This in effect
doubles the voltage swing on the load as compared to a ground referenced load. Plugging twice the voltage
into the power equation, where voltage is squared, yields 4 times the output power from the same supply rail
and load impedance (see equation 1).
V

_ VO(PP)
(rms) -

2.f2

2
V(rms)
Power -~

3-238

(1)

:'I
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. 1B::L
I
GI

aI

102

.:!

102

~

~

Jl0

Y01 +Y02 -

z

-

-

I

1
20

;:::"Y02

r-

100

1k

Y01

Y01 +Yo2

!z

r

J.

'S

Jill

101

>c

==

g

Yo2

~

101

Y01

I

>c

10 k 20k

1
20

100

1k
f - Frequency - Hz

r - Frequency - Hz

Figure 24

Figure 23

~1ExAs

3-258

5.5

103

~ YOO=5Y

I

I

5

OUTPUT NOISE VOLTAGE
va
FREQUENCY

>::L

I

4.5

Figure 22

OUTPUT NOISE VOLTAGE
va
FREQUENCY
103

4

Yoo - Supply Yoltage - Y

INSTRUMENTS
POST OFFICE BOX 656303 • DALLAS. TEXAS 75265

10 k 20k

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOS163B - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS
MAXIMUM PACKAGE POWER DISSIPATION
vs
FREE-AIR TEMPERATURE

POWER DISSIPATION
vs
OUTPUT POWER

0.8
VOO=5V

~

'"

I

c

I

0.6

'iil
III

is

I

0.4

~

~

CD

CI

.=

iE
:I

I

0.75

,g

I

"

0.5

j

:.
I

,p

0.2

0.25

E

~

o

-50

o

-25

25

50

75

o

100

-

RL=8n

c

~

/

t

"'""""-

RL=16n
~

V
o

0.25

TA - Free-Air Temperature - °C

0.5

0.75

1.25

Po - Output Power - W

Figure 25

Figure 26
MAXIMUM OUTPUT POWER
vs
FREE-AIR TEMPERATURE

POWER DISSIPATION
vs
OUTPUT POWER
160

0.5
Voo = 3.3 V

140
~

0.4

oU
I

I

e

c

:I

0

I

RL=8n

0.3

c

I

0.2

I

,p
0.1

o

/

~
V
o

-----

~

0.1

RL=16n

~

120

!

100

E
~

80

i\

8-

i
I

60

RL= 16n

\ ' r--

\ '\

40

t!'

f/

"-

RL=8n

"".

20

0.2

0.4

0.3

0.5

o
o

0.25

0.5

0.75

1.25

1.5

Po - Maximum Output Power - W

Po - Output Power - W

Figure 27

Figure 28

~TEXAS

INSTRUMENTS
POST OFACE 60X 655303 • DALLAS, TEXAS 75265

3-259

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOSI63B - SEPTEMBER 1996 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

vs
LOAD RESISTANCE

vs
SUPPLY VOLTAGE

I AV~-2~N I
f= 1 kHz
CB=0.1I1F
THO+N:!>1%

\

2~--~--~--~----r---~--~

AV=-2VN
f= 1 kHz
CB 0.1 !1F
THD+NS1%
1.51--+---+---+--4---1---::.1

-

it
I
'!5 1.25 1--+--+--t---:::--.l-:--::----p.L-71

1\
\

~

1\

1\

,

"-

I

"- ~=5V

I'..

J"'....r--

""'"

~

t-- ~

VOO=3.3V

f

o
4

8

=

1.75

12

I

I

16 20 24 28 32 36
Load Resistance - n

--

I

2

1---'-I---+~'---I-7"""'-+---I----::""

0.75

0.51--~=--"~--+-~~--I---t

0.25 ~=-:k~~---+--t--+---I

40 44 48

3

Figure 29

vs
FREQUENCY

II

VD~'~'5V
RL=8n
CB=0.1I1F

80

""

Phase
I

f"..I

r'R

Gain

III

0°

I-

-1350
~

-1800

-20

'1M

_2250
10M

1k

10 k

100 k

I

ia

VOO=5V
RL=8n
Bridge-ned Load

-10
-20

~

l
-8001.
= t

o
100

"

,

i\

it\.

10

o

45°

-450

~

~
CI.
CI.
;:,

In
I

r

CB=0.1I1F

I"
'-

-70

.....

II I

~

CB~1 ~

a:: -«l

...i:i

-80
-100
100

f - Frequency - Hz

1k

f - Frequency - Hz

Figure 31

Figure 32

~TEXAS

3-260

5.5

SUPPLY RIPPLE REJECTION RATIO

vs
FREQUENCY

II

5

Figure 30

OPEN-LOOP GAIN

100

3.5
4
4.5
Supply Voltage - V

INSTRUMENTS
POST OFRCE BOX 855303 • DAllAS. TEXAS 75265

10k 20 k

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOS163B - SEPTEMBER 1996 - REVISED MARCH 2000

TYPiCAL CHARACTERiSTiCS
SUPPLY RIPPLE REJECTION RATIO

vs
FREQUENCY

o
-10
-20

-30

............ r-....
............

II III

...~~~I~.~I~F
...............

r--......

"
-60

VDD=5V
RL=sn

-

Single Ended

........

.........

~ ........

7'

CB=1 ~F

-70
-60
-90
-100

100

1k

10k 20k

f - Frequency - Hz

Figure 33

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-261

TPA4861
1-W AUDIO POWER AMPLIFIER
SLOSl63B-SEPTEMBER 1996- REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load versus single-ended mode
Figure 34 shows a linear audio power amplifier (APA) in a bridge-tied load (BTL) configuration. A BTL amplifier
actually consists of two linear amplifiers driving both ends of the load. There are several potential benefits to
this differential drive configuration, but initially, let us consider power to the load. The differential drive to the
speaker means that as one side is slewing up the other side is slewing down and vice versa. This, in effect,
doubles the voltage swing on the load as compared to a ground-referenced load. Plugging twice the voltage
into the power equation, where voltage is squared, yields 4 times the output power from the same supply rail
and load impedance (see equation 1).
V

_ VO(PP)
(rms) 2 Ii
2
V(rms)

(1 )

Power - - -

RL

Voo

Voo

RL

J'

!2X VO(PP)

'V;

-VO(PP)

Figure 34. Bridge-Tied Load Configuration

In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-n speaker from a
singled-ended (SE) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement, which is loudness that
can be heard. In addition to increased power, frequency response is a concern; consider the single-supply SE
configuration shown in Figure 35. A coupling capacitor is required to block the dc offset voltage from reaching
the load. These capacitors can be quite large (approximately 40 IlF to 1000 IlF) so they tend to be expensive,
occupy valuable PCB area, and have the additional drawback of limiting low-frequency performance of the
system. This frequency limiting effect is due to the high pass filter network created with the speaker impedance
and the coupling capacitance and is calculated with equation 2.

~TEXAS

INSTRUMENTS
3-262

POST OFFICE BOX 655303 • DAlLAS. TEXAS 75265

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOS163B - SEPTEMBER 1996 - REVISED MARCH 2000

APPliCATiON iNFORiviATiON
bridged-tied load versus single-ended mode (continued)
f

-

(corner) -

1

21tRL C

(2)

c

For example, a 68-I1F capacitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD

Figure 35. Single-Ended Configuration
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4 times the output power of the
SE configuration. Internal dissipation versus output power is discussed further in the thermal considerations
section.

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the intemal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from Voo.
The internal voltage drop multiplied by the RMS value of the supply current, IOOrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in
the amplifier, the current and voltage waveform shapes must first be understood (see Figure 36).
IDO

,/

-~-

V(LRMS)

IOD(RMS)

Figure 36. Voltage and Current Waveforms for BTL Amplifiers

-!II

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-263

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOSl638-SEPTEMBER 1996- REVISED MARCH 2000

APPLICATION INFORMATION
BTL amplifier efficiency (continued)
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
"different between SE and BTL configurations. In an SE application, the current waveform is a half-wave rectified
shape, whereas in BTL It is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform, both the push and pull transistor are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws currentfrom the supply for halfthe waveform.
The following equations are the basis for calculating amplifier efficiency.
PL

Efficiency = - -

(3)

Psup

Where:

n:V

Efficiency of a· BTL Configuration = 2V P

(4)

DO

Table 1 employs equation 4 to calculate efficiencies for four different output power levels. Note thatthe efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased, resulting
In a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 1. Efficiency Vs Output Power In 5-V 8-0 BTL Systems
Output Power

(W)

Efficiency
(%)

0.25

31.4

Peak·to-Peak
Voltage

(W)

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.S

4.00

0.59

1.25

70.2

4.4rt

0.53

t High peak voltages cause the THO to Increase.

~TEXAS

3-264

Internal
Dissipation

(V)

INSTRUMENTS
POST OFFICE BOX 855303 • DALLAS. TEXAS 75265

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOS163B - SEPTEMBER 1996 - REVISED MARCH 2000

APPLICATION INFORMATION
BTL amplifier efficiency (continued)
A final point to remember about linear amplifiers, whether they are SE or BTL configured, is how to manipulate
the terms in the efficiency equation to utmost advantage when possible. Note that in equation 4, Voo is in the
denominator. This indicates that as Voo goes down, efficiency goes up.
For example, if the 5-V supply is replaced with a 10-V supply (TPA4861 has a maximum recommended Voo
of 5.5 V) in the calculations of Table 1 then efficiency at 1 W would fall to 31"10 and internal power dissipation
would rise to 2.18 W from 0.59 W at 5 V. Then for a stereo 1-W system from a 1O-V supply, the maximum draw
would be almost 6.5 W. Choose the correct supply voltage and speaker impedance for the application.

selection of components
Figure 37 is a schematic diagram of a typical notebook computer application circuit.

50kQ

50kQ

VDD 6

r-~0Ar-~~AAr---------------~1-~--'---VDD=5V

~=

T~

VDot2

Audio
Input

~CI

4 IN-

1W
Internal

3 IN+

Speaker

2

BYPASS

1

SHUTDOWN (see Note A)

V02 8

7

NOTE A. SHUTDOWN must be held low for normal operation and asserted high for shutdown mode.

Figure 37. TPA4861 Typical Notebook Computer Application Circuit

-!I

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-265

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOSl638 - SEPTEMBER 1996 - REVISeD MARCH 2000

APPLICATION INFORMATION
gain setting resistors, RF and RI

The gain for the TPA4861 is set by resistors RF and RI according to equation 5.

G~in

= -

2(~)

(5)

BTL mode operation brings about the factor of 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA4861 is a MOS amplifier, the input impedance is very high;
consequently input leakage currents are not generally a concern, although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values are required for proper startup operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kn and 20 kn. The effective impedance is calculated in equation 6.

R R

Effective Impedance = R F ~
F

+

(6)
I

As an example consider an input resistance of 10 kn and a feedback resistor of 50 kn. The gain of the amplifier
would be -10 VN and the effective impedance at the inverting terminal would be 8.3 kn, which is well within
the recommended range.
For high performance applications metal film resistors are recommended because they tend to have lower noise
levels than carbon resistors. For values of RF above 50 kn the amplifier tends to become unstable due to a pole
formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capaCitor of approximately 5 pF should be placed in parallel with RF This, in effect, creates a low
pass filter network with the cutoff frequency defined in equation 7.

f

1
co(lowpass) - 23tR FC F

(7)

!

For example if RF is 100 kn and Cf is 5 pF then feo is 318 kHz, which is well outside of the audio range.
input capaCitor, CI

In the typical application, an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 8.
1
fco(highpass) = 23tR I CI

(8)

The value of CI is important to consider, as it directly affects the bass (lOW frequency) performance of the circuit.
Consider the example where RI is 10 kn and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.

C -

1

(9)

I - 23tR,fco

In this example, C, is 0.40 IlF, so one would likely choose a value in the range of 0.47 IlF to 1 IlF. A further
consideration for this capacitor is the leakage path from the input source through the input network (R" C,) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voltage atthe inputto the amplifier
that reduces useful headroom; especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capaCitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. Please note that it is important to confirm the capaCitor polarity in the application.

~TEXAS

3-266

INSTRUMENTS
POST OFRCE BOX 656303 • DALLAS. TEXAS 75265

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOSl63B- SEPTEMBER 1996- REVISED MARCH 2000

AppliCATiON iNfORiiiiATiON
power supply decoupling, Cs
The TPA4861 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure that the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 IlF placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytiC capacitor of 10 IlF or greater placed near
the power amplifier is recommended.

midrall bypass capacitor, CB
The midrail bypass capacitor, Ce, serves several important functions. During start-up or recovery from
shutdown mode, Cs determines the rate at which the amplifier starts up. This helps to push the start-up pop
noise into the subaudible range (so slow it can not be heard). The second function is to reduce noise produced
by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation
circuit internal to the amplifier. The capacitor is fed from a 25-kn source inside the amplifier. To keep the start-up
pop as low as pOSSible, the relationship shown in equation 10 should be maintained.

1
(C s x 25

kn)

s_1_
(CIR I)

(10)

As an example, consider a circuit where Cs is 0.1 IlF, CI is 0.221lF and RI is 10 kO. Inserting these values into
the equation 9 we get:
400 s 454
which satisfies the rule. Bypass capacitor, Cs, values of 0.11lF to 11lF ceramic or tantalum low-ESR capacitors
are recommended for the best THO and noise performance.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-267

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOS163B - SEPTEMBER 1996 - REVISED MARCH 2000

APPLICATION INFORMATION

single-ended operation
Figure 38 is a schematic diagram of the recommended SE configuration. In SE mode configurations, the load
should be driven from the primary amplifier output (V01, terminalS).

voo 6

.r

RF
Audio

Vo1)/2

Input

~c~

Voo

±
i1rJ

Cs

-=-

RI

4

,A

IN-

I

I
3

IN+

V01 5

r+
'v-

J

250-mW
External

Speaker

-

CBt

-=2

BYPASS

+ 1

V(J2 8

RSE=500

+

CSE=O.1 IlF ±

Figure 38. Singled-Ended Mode
Gain is set by the RF and RI resistors and is shown in equation 11. Since the inverting amplifier is not used to
mirror the voltage swing on the load, the factor of 2 is not included.
Gain = -

(~~)

(11)

The phase margin of the inverting amplifier into an open circuit is not adequate to ensure stability, so a
termination load should be connected to Vo2. This consists of a 50-0 resistor in series with a 0.1-IlF capacitor
to ground. It is important to avoid oscillation of the inverting output to minimize noise and power dissipation.
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the following relationship:
1
(C e x 25

<_1_~_1_

kn) -

(CIR I)

RLC C

~TEXAS

3-268

INSTRUMENTS
POST OFRCE SOX 655303 • DALLAS, TEXAS 75265

(12)

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOSl63B - SEPTEMBER 1996 - REVISED MARCH 2000

APPLiCATiON iNFORMATiON

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cd is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 13.

f

.
=
1
outh 19h
23tR L C c

(13)

The main disadvantage, from a performance standpoint, is that the load impedances are typically small, which
drives the low-frequency comer higher. Large values of Cc are required to pass low frequencies into the load.
Consider the example where a Cc of 68 IlF is chosen and loads vary from 8 n, 32 0, and 47 kQ. Table 2
summarizes the frequency response characteristics of each configuration.

Table 2. Common Load Impedances vs Low Frequency Output Characteristics In SE Mode

RL

Cc

Lowest Frequency

SO

68J.lF

293 Hz

320

6SJ.lF

73Hz

47,0000

6SJ.lF

0.05 Hz

As Table 2 indicates, most of the bass response is attenuated into 8-0 loads, while headphone response is
adequate and drive into line level inputs (a home stereo for example) is very good.

shutdown mode
The TPA4861 employs a shutdown mode of operation designed to reduce supply current, IOD(q) , to the absolute
minimum level during periods of nonuse for battery-power conservation. For example, dUring device sleep
modes or when other audio-drive currents are used (Le., headphone mode), the speaker drive is not required.
The SHUTDOWN input terminal should be held low during normal operation when the amplifier is in use. Pulling
SHUTDOWN high causes the outputs to mute and the amplifier to enter a low-current state,
IOO(SO) = 0.6 !lA. SHUTDOWN should never be left unconnected because amplifier operation would be
unpredictable.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real capacitor can be modeled
simply as a resistor in series with an ideal capacitor. The voltage drop across this resistor minimizes the
beneficial effects of the capacitor in the circuit. The lower the equivalent value of this resistance the more the
real capacitor behaves like an ideal capacitor.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-269

TPA4861
1·W AUDIO POWER AMPLIFIER
SLOSl63B - SEPTEMBER 1996 - REVISED MARCH 2000

APPLICATION INFORMATION

thermal considerations
A prime consideration when designing an audio amplifier circuit is internal power dissipation in the device. The
curve in Figure 39 provides an easy way to determine what output power can be expected out of the TPA4861
for a given system ambient temperature in designs using 5-V supplies. This curve assumes no forced airflow
or additional heat sinking.
160

I

Voo=5V
140

oU
I

120

I!!

I&

100

..

80

E
(!

1
I

60
40

RL=160

\.

\'

~

\ '\

""

i'...

RL=80

.".

~
20

o
o

0.25

0.5

0.75

1.25

Po - Maximum Output Power - w

Figure 39. Free-Air Temperature vs Maximum Continuous Output Power

s-y versus 3.3-Y operation
The TPA4861 was designed for operation over a supply range of 2.7 V to 5.5 V. This data sheet provides full
speCifications for 5-V and 3.3-V operation, as these are considered to be the two most common standard
voltages. There are no special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain
setting, or stability. Supply current is slightly reduced from 3.5 mA (typical) to 2.5 mA (typical). The most
important consideration is that of output power. Each amplifier in TPA4861 can produce a maximum voltage
swing of Voo - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as opposed
to when VO(PP) = 4 V while operating at 5 V. The reduced voltage swing subsequently reduces maximum output
power into an 8-0 load to less than 0.33 W before distortion begins to become significant.
Operation at 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds of the supply power for a given output-power level than operation from 5-V supplies.
When the application demands less than 500 mW, 3.3-V operation should be strongly considered, especially
in battery-powered applications.

~TEXAS

3-270

INSTRUMENTS
POST OFFICE BOX 665303 • DALLAS. TEXAS 75285

TPA0253
1-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S280B - JANUARY 2000 - REVISED MARCH 2000

DGQPACKAGE

• Ideal for Notebook Computers, PDAs, and
Other Small Portable Audio Devices
• 1 W Into 8-0 From 5-V Supply

(TOP VIEW)

FILT-CAP
SHUTDOWN

• 0.3 W Into 8-0 From 3-V Supply
• Stereo Head Phone Drive
• Mono (BTL) Signal Created by Summing
Left and Right Signals Internally

LO/MOLIN
GND
STIMN
ROIMO+

• Wide Power Supply Compatibility
2.5 Vto 5.5 V
• Low Supply Current
- 3.2 mA Typical at 5 V
- 2.7 mA Typical at 3 V
• Shutdown Control •.• 1 !LA Typical
• Shutdown Pin is TTL Compatible
• -40°C to 85°C Operating Temperature
Range
• Space-Saving, Thermally-Enhanced MSOP
Packaging
description
The TPA0253 is a 1-W mono bridge-tied-Ioad (BTL) amplifier designed to drive speakers with as low as 8-0
impedance. The mono signal is created by summing left and right inputs internally. The amplifier can be
reconfigured on-the-fly to drive two stereo single-ended (SE) signals into head phones. This makes the device
ideal for use in small notebook computers, PDAs, digital personal audio players, anyplace a mono speaker and
stereo head phones are required. From a 5-V supply, the TPA0253 can delivery 1-W of power into a 8-0 speaker.
The gain of the input stage is set by the user-selected input resistor and a 50-kQ internal feedback resistor
(AV =- RF/ RI)· The power stage is internally configured with a gain of -1.25 VN in SE mode, and -2.5 VN in
BTL mode. Thus, the overall gain of the amplifier is 62.5 kQf RI in SE mode and 125 kQf RI in BTL mode. The
input terminals are high-impedance CMOS inputs, and can be used as summing nodes.
The TPA0253 is available in the 10-pin thermally-enhanced MSOP package (DGQ) and operates over an
ambient temperature range of -40°C to 85°C.

A..

~

Please be aware that an important notice concerning availability. standard warranty. and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments InCOrporated.

~TEXAS

Copyright © 2000. Texas Instruments Incorporated

INSTRUMENTS
POST OFFICE BOX 655303 • OAUAS. TEXAS 75265

3-271

TPA0253
1·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS280B - JANUARY 2000 - REVISED MARCH 2000

4

,------------Voo

3

1

VOO

I
1I
,---:...J F1LT-CAP
-L
I

I

BYPASS

Voo

50kn

iI

50kn
1.25*R

I
51 RIN
Right
Audio

Input

CI
II--",R",11r-'
BYPASS

50kn
StereolMono
Control

50kn
50kn

STIMN

1.25*R

Left
Audio CI

Input

9 UN
Ir-~R~I~~---*--~

From
System Control

BYPASS

Shutdown
and Depop
Circuitry

L _________________________

~TEXAS

INSTRUMENTS
3-272

-=-

Cc

LOIMO- 1 10

I
I
I
I
I
21 SHUTDOWN

I
I
I
I
I
I
I7
I
I
I
I
I
I

POST OFRCE BOX 655303 • DAUAS, TEXAS 75265

I
I
I
I
I
I
I
I
I
I

~

1 kO

TPA0253
1·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S280B - JANUARY 2000 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

MSOP't
(DGQ)

-40°C to 85°C

TPA0253DGQ

MSOP
SYMBOLIZATION
AEL

t The DGQ package are available taped and reeled. To order a taped and reeled part, add the
suffix R to the part number (e.g., TPA0253DGQR).

Terminal Functions
TERMINAL
NAME
NO.

I/O

DESCRIPTION

FILT-CAP

1

SHUTDOWN

2

I

Terminal used to filter power supply

VDD

3

I

Positive power supply

BYPASS

4

I

Mid-rail bias voltage

TTL-compatible shutdown terminal

RIN

5

I

Right-<:hannel input terminal

ROIMO+

6

0

Right-output in SE mode and mono positive output in BTL mode

STIMN

7

I

Selects between Stereo and Mono mode. When held high, the amplifier is in SE stereo mode, while held
low, the amplifier is in BTL mono mode.

GND

8

LIN

9

I

Left-<:hannel input terminal

LOIMQ-

10

0

Left-oulput In SE mode and mono negative output in BTL mode.

Ground terminal

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1116 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions' is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
DGQ

DERATING FACTOR
2.14~

17.1 mW/"C

1.37W

1.11 W

11 Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-273

TPA0253
1·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS280B - JANUARY 2000 - REVISED MARCH 2000

recommended operating conditions
Supply voltage, VOO
STIMN

High-level input voltage, VIH

MAX

2.5

5.5

I VOO=3V

2.7

IVOO=5V

4.5

SHUTOOWN

V

V

1.65
V

2.75

SHUTOOWN

0.8

-40

Operating lree-air temperature, TA

electrical characteristics at specified free-air temperature, VDD
noted)
PARAMETER

UNIT

2

IVOO=3V
I VOO=5V

STIMN

Low-level input voltage, VIL

MIN

85

°C

=3 V, TA =25°C (unless otherwise

TEST CONOITIONS

IVool

Output offset voltage (measured differentially)

VIO=O.I%,

Gain=8dB

PSRR

Power supply rejection ratio

VOO=2.9Vt03.1 V,

BTL mode

IIIHI

High-level input current

VOO=3.3V,

VI=VOD

VOO=3.3V,

VI=O

MIN

TYP

MAX

UNIT

30

mV

1

jIA

65

dB

IIILI

Low-level input current

ZI

Input impedance

50

100

Supply current

2.7

4

rnA

IOO(SO)

Supply current, shutdown mode

1

10

jIA

operating characteristics, VDD

TEST CONDITIONS
BTL mode,

Gain = 14dB

THO=O.I%
Gain = 1.9 dB

SEmode,

RL=320

Output power, see Note 1

THO+N

Total harmonic distortion plus noise

Po=250mW,

1= 20 Hz to 20 kHz

80M

Maximum output power bandwidth

Gain = 1.9 dB,

THO =2%

NOise output voltage

MIN

THO=O.I%,
Po

Vn

1= 1 kHz,

CB = 0.47 I1F,

CB = 0.47 I1F

1=20 Hz to 20 kHz

NOTE 1: Output power is measured at the output terminals 01 the device at I = 1 kHz.

~TEXAS

INSTRUMENTS
3-274

jIA

kn

=3 V, TA =25°C, RL =4 n, f =1 kHz (unless otherwise noted)

PARAMETER

Supple ripple rejection ratio

1

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TYP

MAX

UNIT

300
30

mW

0.2%
20
BTL mode

46

SEmode

68

BTL mode

83

SEmode

33

kHz
dB

I1V RMS

TPA0253
1-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S280B - JANUARY 2000 - REViseD MARCH 2000

electrical characteristics at specified free-air temperature, Voo = 5 V, TA = 25°C (UnleSS otherwise
noted)
,
TEST CONDITIONS

PARAMETER
IVool

Output offset voltage (measured differentially)

VIO=O,

Gain=8dB

PSRR

Power supply rejection ratio

VOO=4.9Vto5.1 V,

BTL mode

IIIHI

High-level input current

VOO=5.5V,

VI=VOO

VOO=5.5V,

VI=O

MIN

TYP

MAX

UNIT

30

mV

1
1

!LA
!LA

62

dB

IIILI

Low-level input current

ZI

Input impedance

50

100

Supply current

3.2

4.8

mA

IOO(SO)

Supply current, shutdown mode

1

10

!LA

operating characteristics, Voo

=5 V, TA =25°C, RL =4 0., f =1 kHz (unless otherwise noted)

PARAMETER

TEST CONDITIONS
THO =0.1%,

BTL mode

THO =0.1%,

SEmode,

Po

Output power, see Note 1

THO+N

Total harmonic distortion plus
noise

PO=1W,

f = 20 Hz to 20 kHz

BOM

Maximum output power bandwidth

Galn=8dB,

THD=2%

Supple ripple rejection ratio

1=1 kHz,

CB =0.47ILF

Noise output voltage

CB=0.47ILF,

1= 20 Hz to 20 kHz

Vn

kn

MIN

TYP
1

RL=32Q

85

MAX

UNIT
W
mW

0.33%

20
BTL mode

46

SEmode

60

BTL mode

85

SEmode

34

kHz
dB

ILVRMS

NOTE 1: Output power IS measured at the output terminals 01 the deVice at 1= 1 kHz.

:ilTEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-275

3-276

TPA010a
1.7S-W a-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997- REVISED MARCH 2000

• Desktop Computer Amplifier Solution
- 1.75-W Bridge Tied Load (BTL) Center
Channel
- 500-mW LlR Single-Ended Channels

PWPPACKAGE
(TOP VIEW)

• Low Distortion Output
- < 0.05% THD+N at Full Power
• Full 3.3-V and SOV Specifications
• Surface-Mount Power Package
24-Pin TSSOP
• LlR Input MUX Feature
• Shutdown Control ••• 100

VDD
SHUTDOWN
MUTE OUT
COUT+
MODEB
GNDIHS

=51JA

24
23
22
21
20
19
18
17
16
15
14

10
2
3
4
5
6
7

GNDIHS
NC
LOUT
LLiNEIN
LHPIN
CIN

8
9
10
11
12

GNDIHS
NC
ROUT
RLiNEIN
RHPIN
BYPASS
VDD
NC
HP/LINE
COUTMODE A
GND/HS

13

CFCt RFC
6

•

C

19

RILC
CBT

9

NC

-=-

8

ClN

MUTE OUT

---1CII

RIR

20

RHPIN

21

RLiNEIN

-/

COUT- 15

I.......

I

RIL

5
4

IL
RFR

RFL

LHPIN
LLiNEIN

~MODEB

11

Speaker

'v

RM2

VDD

VDD

RM1

VDD 718 V
DD
HP/LINE 16

~

r---

~

Right
MUX

ROUT 22

COUTR
If
1\

+

'-

?

<
~

...L

~

Left
MUX

-

InternaI'

MODE A 14

T

NC

---1C

+

JV
-

IR

I
I

~

SHUTDOWN

-1NC

r-

BYPASS

COUT+ 10

RM3

-=
LOUT 3

+

-=I,

I'
CoUTL

GND/HS
~ 1, 12, 13, 24

.,

~

Please be aware that an Important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS

Copyright ill> 2000, Texas Instruments Incorporated

INSTRUMENTS
POST OFFICE BOX 65S303 • DA1.LAS. TEXAS 75265

3-277

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

description
The TPA0103 is a 3-channel audio power amplifier in a 24-pin TSSOP thermal package primarily targeted at
desktop PC or notebook applications. The left/right (LlR) channel outputs are single ended (SE) and capable
of delivering SOO mW of continuous RMS power per channel into 4-Q loads. The center channel output is a
bridged tied load (BTL) configuration for delivering maximum output power from PC power supplies. Combining
the SE line drivers and high power center channel amplifiers in a single TSSOP package simplifies design and
frees up board space for other features. Full power distortion levels of less than 0.2S% THD+N into 4-Q loads
from a S-V supply voltage are typical. Low-voltage application are also well served by the TPA01 03 providing
800 mW to the center channel into 4-Q loads with a 3.3-V supply voltage.
Amplifier gain is externally configured by means of two resistors per input channel and does not require external
compensation for settings of 1 to 10. A two channel input MUX circuit is integrated on the LlR channel inputs
to allow two sets of stereo inputs to the amplifier. In the typical application, the center channel amplifier is driven
from a mix of the LlR inputs to produce a monaural representation of the stereo signal. The center channel
amplifier can be shut down independently of the LlR output for speaker muting in headphone applications. The
TPA0103 also features a full shutdown function for power sensitive applications holding the bias current
to S!lA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of less than 3SoC/W are readily realized in multilayer PCB
applications. This allows the TPA0103 to operate at full power at ambient temperature of up to 8SoC.
AVAILABLE OPTIONS
PACKAGE
TA

TSsopt

-40°C to 85°C

TPA0103PWP

(PWP)

tThe PWP package is available in left·ended tape
and reel only (e.g., TPA0103PWPLE).

~TEXAS

INSTRUMENTS
3-278

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME

NO.

110

DESCRIPTION

BYPASS

19

CIN

6

I

Center channel input

COUT+

10

0

Center channel + output. COUT+ is in an active or high-impedance state unless the device is in a mute state
when the MODE A terminal (14) is high and the MODE B terminal (11) is low.

COUT-

15

0

Center channel - output. COUT- is in an active or high-impedance state unless the device is in a mute state
when the MODE A terminal (14) is high and the MODE B terminal (11) is low.

GNDIHS

1,12,
13,24

MODE A,
MODEB

14,11

Bypass. BYPASS is a tap to the voltage divider for the intemal mid-supply bias.

Ground. GND/HS is the ground connection for circuitry, directly connected to thermal pad.
I

Mode select. MODE A and MODE B determine the output modes of the TPAOI 03.
TERMINAL

3 CHANNEL

MUTE

MODE A

L

H

L

H

MODEB

L

L

H

H

CENTER
ONLY

UR
ONLY

HP/LINE

16

I

LHPIN

5

I

Left channel headphone input, selected when the HPILINE terminal (t 6) is held high

LLiNEIN

4

I

Left channel line input, selected when the HPILINE terminal (16) is held low

LOUT

3

0

Left channel output. LOUT is active when the MODE A terminal (14) is low and the MODE B terminal (11) is
don't care.

MUTE OUT

9

0

When the MODE A terminal (14) is high and the MODE B terminal (11) is low, MUTE OUTis high and the device
is in a mute state. Otherwise MUTE OUT is low.

NC

2,17,
23

Input MUX control input, hold high to select (UR) HPIN (5, 20), hold low to select (UR) LlNEIN (4, 21). HP/LINE
is normally connected to ground when inputs are connected to (UR) LlNEIN.

No intemal connection

RHPIN

20

I

Right channel headphone input, selected when the HP/LINE terminal (16) is held high

RLINEIN

21

I

Right channel line input, selected when the HPILINE terminal (16) is held low

ROUT

22

0

Right channel output. ROUT is active when the MODE A terminal (14) is low and the MODE B terminal (11)
is don't care.

8
7,18

I

Places entire IC in shutdown mode when held high, 100 5 !1A

I

Supply voltage input. The VDD terminals must be connected together.

SHUTDOWN
VDD

=

~TEXAS

INSTRUMENTS
POST OFFICE eox 65s303 • DALLAS. TEXAS 75265

3-279

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A- JULY 1997 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Continuous output current (COUT+, COUT-, LOUT, ROUT) ...............................•..... 2 A
Continuous total power dissipation ................................................ internally limited
Operating virtual junction temperature range, TJ ........................•........... -40°C to 150°C
Operating virtual case temperature range, Tc ...................................... -40°C to 125°C
Storage temperature range, Tstg ....................................... ...... ...... -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds
.............................. 260°C

t Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions' is not
Implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

DERATING FACTOR
2.7W

1.7W

21.SmWrC

1.4W

:I: Please see the Texas Instruments document. PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more Information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
MIN

NOM

MAX

3

5

5.5

Supply Voltage, VOO
Operating junction temperature, TJ

I

UNIT
V
·C

125

dc electrical characteristics, TA = 25°C
PARAMETER
VOO=5V
100

TVP

MAX

3 Channel

19

25

mA

Land R or Center only

9
13

15
20

rnA
rnA

TEST CONDITIONS

Supply current
VOO= 3.3 V

3 Channel

NOM

Land R or Center only

3

10

rnA

Gain =2,

5

35

mV

15

IIA
IIA

Voo

Output offset voltage (measured differentially)

VOO=5V,

IOOIMUTE)

Supply current in mute mode

VOO=5V

SOO

100 in shutdown
VOO=5V
IOO(SO)
NOTE 1: At 3 V < VOO < 5 V the de output voltage is approximately Vo0f2.

5

~TEXAS

3-280

UNIT

INSTRUMENTS
POST OFFICE BOX 6S5303 • OAUAS, TEXAS 75285

See Note 1

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997- REVISED MARCH 2000

ac operating characteristics, VDD = 5 V, TA = 25°C, RL :: 4 n
PARAMETER

Po

TEST CONDITIONS

Output power (each channel) (see Note 2)

MIN

BTL,

Center channel

1.75

THD=1%,

BTL,

Center channel

2.1

THD = 0.2%,

SE,

LJR channels

535

THD=1%,

SE,

LJR channels

575

THD+N

Total harmonic distortion plus noise

Po= 1.5W,

1= 20 to 20 kHz

BOM

Maximum output power bandwidth

G=10,

THD<5%

Phase margin

Open loop

Supply ripple rejection ratio
1=20-20kHz

W

mW

kHz

>20

0

Center channel

80

LJR channels

58

Center channel

60

LJR channels

30

dB

65

Channel-to-channel output separation

1 = 1 kHz

LinelHP input separation
Input impedance
Signal-to-noise ratio

VO= 1 V(rms)

Output noise voltage

BTL,

Center channel

SE,

LJR channels

BTL,

Center channel

SE,

LJR channels

UNIT

0.25%

Mute attenuation

Vn

MAX

85

1 = 1 kHz

ZI

TYP

THD=0.2%,

dB

95

dB

100

dB

2

MQ

94

dB

100
20

l1V (rms)

9

NOTE 2: Output power is measured at the output terminals 01 the IC at 1 kHz.

ac operating characteristics, VDD = 3.3 V, TA = 25°C, RL = 4 n
PARAMETER

TEST CONDmONS
THD = 0.2%

BTL,

THO = 1%
THD=0.2%,

BTL,

Center channel

650

SE,

LJR channels

215

LJR channels

Output power (each channel) (see Note 2)

THD = 1%,

SE,

THD+N

Total harmonic distortion plus noise

Po =750mW,

1 = 20 to 20 kHz

BOM

Maximum output power bandwidth

G=10,

THD<5%

Phase margin

Open loop

Supply ripple rejection ratio
1=20-20kHz

ZI

70
62

Center channel

55

LJR channels

30

Input impedance
Signal-to-noise ratio

Vn

Output noise voltage

VO= 1 V(rms)

>20

LJR channels

LinelHP input separation

BTL,

Center channel

SE,

LJR channels

UNIT

mW

235

Center channel

1 = 1 kHz

MAX

0.8%

85

Mute attenuation
Channel-to-channel output separation

TYP
800

Po

f = 1 kHz

MIN

Center channel

kHz
0

dB

85

dB

95

dB

100

dB

2

MQ

93
100

BTL,

Center channel

21

SE,

LJR channels

10

dB

l1V(rms)

NOTE 2: Output power is measured at the output terminals 01 the IC at 1 kHz.

-!I1TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-281

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION
RF

-1

RI
CB

CI

4.7 11F

RL=4QorSQ

T
-=-

-::-

VDD

-=MODE A
MODEB
SHUTDOWN

Figure 1. BTL Test Circuit

CB
4.7j1f

T

MODE A
MODEB

VDD
VDD

VDD

SHUTDOWN
-::-::-

RF

--1

CI

Co
RI

1~

-::-

--1

CJ

Co
RI

1"'

RF

Figure 2. SE Test Circuit

~1ExAs

3-282

INSTRUMENTS
POST OFFICE BOX 656303 • DAl.I..AS, TEXAS 75285

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A- JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE

Output noise voltage

vs Frequency

Supply ripple rejection ratio

vs Frequency

Crosstalk

vs Frequency

Open loop response

vs Frequency

3,4,7,10-12,15,18,21,24,
27,30,33,36
5,6,8,9,13,14,16,17,19,
20, 22, 23, 25, 26, 28, 29, 31,
32,34,35
37,38
39,40
41,42
43,44

Closed loop response

vs Frequency

45-46

Supply current

vs Supply voltage

Po

Output power

Supply voltage
vs Load resistance

Po

Power dissipation

vs Output power

49
50,51
52,53
54-57

vs Output power
THO+N

Total harmonic distortion plus noise
vs Frequency

Vn

100

~

VB

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10

=
VOO=5V
- f=1kHz

I

Iz

-

I

BTL

+

c

~

I

I

II

I

~

..

~
:I:

C

:!

t--

_ RL=Sn

0.1

~

Z

+

"'"'"

Q

0.01

o

......

I

'\.

Q

i!:

0.25 0.5 0.75 1 1.25 1.5 1.75

2 2.25 2.5

I
I
RL=4n

~

~I

If

I

I

!

!

i!:

I

I
I

~

~

J

0.1

••+
Z

.2
c

RL=4n
RL=sn

~ VOO=5V

r- f= 1 kHz
r- SE

i

is
'j!
0

~
I

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10
r-

0.01

o

==

-

75 150 225 300 375 450 525 600 675 750
Po - OUtput Power - mW

Po - Output Power - W

Figure 3

Figure 4

-!!1 TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

3-283

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

fI!.

VOO=5V
PO=1.5W
RL=4n
BTl

I

I

fI!.

+
c

8

~

!

1
:z:

,~

AV =-20 VN

AV=-10VN

V i-'"

"'"

0.1

I
I

Z

~
V

100

1k
f - Frequency - Hz

PO=1.5W

.~

PO=0.75W -

0

E
:!
I

Po = 0.25 W

II

0.01

100

20

vs

OUTPUT POWER

FREQUENCY

fI!.

VOO=5V
RL=4n
BTl

VOO=5V
RL=sn
AV=-2VN
BTL

·S•
z

+
c

~

0

'E

I

~

f: 20 kHz

Q

:!

10

I

+
c

1

.!i

c

~

0.1

I

0

..E
:z:

1= 1 kHz

PO=0.5
0.1

!

~

I

f=20Hz

j!:

I IIII

0
0.01
0.01

Z

......

0
:z:

I-

0.01

0.1
Po - Output Power - W

10

20

Figure 7

II

I II

II

II

100

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

V

PO =0.25W

-

I I IIIIIII

1k
f - Frequency - Hz

FigureS

W~

.,

PO=1W

I

Z

3-284

10 k20 k

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

I

1k
f - Frequency - Hz

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

I

1111

II IIII

I-

Figure 5

fI!.

~

0.1

Z

10 k 20k

/A

\

0
:z:

II IIII

0.01
20

~

I

AV= 2VN -

1'11111111 -

0
:z:

I-

VOD=5V
RL=4n
AV =-2 VN
BTL

·ftZ

+

1\1

10

I

10k 20k

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

'#.

vs

vs

FREQUENCY

OUTPUT POWER

10

'#.

VOO=5V
PO=1W
RL=8D
BTL

I

Iz

YOO=5Y
RL=8D
Ay =-2 VN
BTL

Iz

+
c

+
c

~

I
~
I!

10

I

~

0

0.1

Ay = -20 YN

Ay=-10YN

........

......

;;;;3
/

g

I
~
I!

f= 20 kHz

r--.

0

0.1

g

I

f=1 kHz

I

Ay=-2YN -

Z

+
Q
::c

Z

+
Q
::c

11111111

t-

0.01
20

100

f=20Hz

0.1
Po - OUtput Power - W

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10

I

~ VOo=3.3Y

'#.

I- BTL

.I!0

z

I

+
c

I

~

I

.!!
c

"

~ VOO=3.3Y
~f=1kHz
I- SE

z

I
t= 1==

RL=4D
II

RL=8D

I

~

I
I

I-- -

RL=8D

0

Ii!!
0.1

~

I

~

I

I

+
c

.!!
c

0

!

10

I

~ f=1 kHz

.I0

10

Figure 10

Figure 9

'#.

I

111111

0.01
0.01

10k 20 k

1k

~

t-

111111

I

~

Z

Z

0.1

i

I

:t

I

\..

I

.:!i

RL=4D -

.:!i

~

~

0.01

o

0.01
0.1

0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
Po - Output Power '- W

1

o

30

Figure 11

60

90 120 150 180 210 240 270 300
Po - OUtput Power - mW

Figure 12

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-285

TPA0103
1.7S-W 3-CHANNELSTEREO AUDIO POWER AMPLIFIER
SLOSl67A - JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

il-

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY

FREQUENCY

vs

10

Jl

~
+
c

I
+
c

~

i

~0
IE

j

"

0.1

Ay=-20YN

_100-

, I IL

.....

Ay=-10YN

----

i

I
~

I

c
0

II ill

j!:

I 1k

0.01
20

100

......~

0.1

PO=O.l W

i

I

+

I II

I

j!:
0.01
10k 20k

100

f - Frequency - Hz

Figure 13

Figure 14

20

"'
II I

10k 20k

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
OUTPUT POWER

FREQUENCY

vs

10

!z

+
c

..........

0

il-

YOO = 3.3 Y
RL=4Q
Ay =-2 YN
BTL

..... 1'-0

I

I

~0

0.1

-... ..... 1"-

I

.:1

1=1 kHz

0.01
0.01

BTL

i

~I

i

11111

j!:

RL=8Q

I
I

f=20kHz

1=20 Hz

Z

YOO=3.3Y
10~~~
Po = 0.4W

+

J
IIII

0.1
Po - output Power - W

10
f - Frequency - Hz

Figure 15

Figure 16

~1EXAS

3-286

-

Po = 0.35 W

Z
Q

lk
f - Frequency - Hz

I

j

PO=0.7~

IE

!

Ay =-2 YN

.:1

I

..r

.2

~

Z

il-

YOO = 3.3 Y
RL=4Q
Ay=-2YN
BTL

I

~

:!

10

il-

YOO=3.3Y
Po = 0.75W
RL=4Q
BTL

I

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10
il-

~
III-

I

110
z

+

il-

VOO=3.3V
RL=8U
AV=-2VN
BTL

10
VOO=3.3V
RL=8U
AV=-2VN
BTL

..
I

II

"0

z

+

c

c

~

i

,

~0
I§

fI7

:I!

0.1

~

~
......

.~
0

:I:

OJ

PO=OAW

0.1

Z

Z

j!:

:I:

~

+

Q

POI~,0.1 W

0.01
20

100

I-

1k

........

f= 1 kHz

-

~I

I

f= 20 kHz

is

i

/~

PO-O.25W

!

......

~

III

f

11111
0.1

0.01
0.01

10k 20k

f - Frequency - Hz

10

F

vs

FREQUENCY

FREQUENCY
10

Voo=5V

ilII

I- RL=4U
I- SE

·6z

~

~

~0
I§

r--

r0.1

A

Av= 10VN

111111

V

V

1

AV =-5 VN

=~

is

.~

, ........
PO=0.5W

:I:

.,.

I

0

.I§

!

~

I

1111111

0

'E

i

:I:

SE

+
c

c

!

VOO=5V
RL=4U
AV= 2VN

I

+

..

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

~ PO=0.5W

iz

10

Figure 18

TOTAL HARMONIC DISTORTION PLUS NOISE

I

I

Po - Output Power - W

Figure 17

il-

~'20 Hz

/.

I I
Po = 0.25W

0.1

~

I

I

AV=-1 VN

Z

+
Q

j!:
0.01
20

100

111111

I I I

I IIII

I

1k
f - Frequency - Hz

"

Z

+
Q

j!:

Po =0.1 W ......
1"1 IIIIII

0.01
10k 20 k

20

100

-

.fo'
1k

10 k 20 k

f - Frequency - Hz

Figure 19

Figure 20

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-287

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
FREQUENCY

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
OUTPUT POWER
~

10

Iz

+

~
I

VOO=5V
RL=4C
AV =-2 VN
SE

I

SE

+

i

I

1= 20kHz

-- ....

.~

~

!

i== VOD=5V

r- Po=O.25W
r- RL=8C

I

l""- t--..

c

~

10

I
!

0.1

~

f=100Hz

1i
~

-

AV=-10VN
...

AV=~VN

~

+
Q

:z:
I-

AV=-1 VN
ill

j!:

1=1 kHz

I Jill

0.01

0.01
0.001

20

0.01
0.1
Po - Output Power - W

100

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

10
~

VDD=5V
RL=8C
SE

Jl0

10

I

+
c

t- SE

t-

+

.2
"C
0

~

.S:!
c

~

0

~

0.1

~

Po = 0.25 W

~I

+
Q

~O=0.1W

j!:

11'H-:~.-""I'"

0.01

20

I

..;.i

Z

Il~

~

Po = 0.05 W

100

I
I

1k

10k 20k

1= VOo=5V

t- RL=8C
t- AV=-2VN

I

z

Z

1= 20 kHz

r....
0.1
~

1=1 kHz
I.....

~

0.01
0.001

LIL

/'

T,l1

1= 100 Hz

f - Frequency - Hz

0.01

0.1

Po - Output Power - W

Figure 23

Figure 24

~TEXAS

3-288

10k 20k

Figure 22

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
FREQUENCY
I

I

1k
f - Frequency - Hz

Figure 21

I1i

L

11111

0.1

I

I

Z

~

L

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997 - REVISED MARCH 2000

TfFiCAi. CHARACTERiSTiCS

'#.

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY

FREQUENCY

10

vs

F VOO=5V

'#.

~ Po=75mW

I

110

i

+

+
c

c

~

i

~0

~

AV=~VN

0.1

~0

..
~

III

::t:

~

j!:

20

100

j!:

f - Frequency - Hz

1k
f - Frequency - Hz

Figure 25

Figure 26

100

20

10k 20k

TOTAL HARMONIC DISTORTION PLUS NOISE
FREQUENCY

vs

10

'#.

=

:: VOO=5V
RL=32n
_ SE

10

I

....

....:

+

c
0

't:

~

~

0

~

!

-- -..

I

Z

f=20Hz

Z

f=2OkHz

0.1

~I

t:"---.
........ .....

0.01
0.001

~

0

-

~0

VOO = 3.3 V
PO=0.2W
RL=4n
SE

I

0

...

VpO=25 mW
0.01

10k 20 k

1k

't:

::t:

~

lA'11

0

+
c

+
C

Po=75mW

vs
OUTPUT POWER

I
~

Po=50mW

TOTAL HARMONIC DISTORTION PLUS NOISE

I

::t:

=

Z

I 1111

0.01

A

0.1

I

II~YI~-1 ~~

Z
+
C

'#.

I

Av=-10VN

E

!
~I

VOO=5V
RL=32n
SE

11

... RL=32n
r- SE

z

10

I

lill

!

/

AV=-10VN

"

I lLll-0.1

AV=~VN

~
I

IYJl f =1kHz

~ Iffi>..fflllllil

I

AV=-1 VN

+
C
::t:

I II ill

...

TmIT

0.01

0.01
0.1
Po - Output Power - W

I I

20

Figure 27

100

1k
f - Frequency - Hz

10k 20k

Figure 28

~lExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-289

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

vs

FREQUENCY

OUTPUT POWER

10

~

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

=

Voo =3.3 V
RL=40
SE

I

Iz

:
-

+

10
I
III

.!I

z0

r-....

+

I:
0

~~

i:

i

i:
0
/'/

Po = 0.1 W

E
!

....

I:
0

PO=0.2W

~0

VOO=3.3V
RL=40
Av =-2 VN
SE

"1ft.

V

/~

0.1

~

f=2OkHz

~0

~
:z:

r-...

II

....

f=1 kHz

0.1

!

J

~

I

I

Z

j!:
0.01
20

Z

PO=0.05W

-

+
Q

+

11111

100

1k

:z:

I-

"

10k 20 k

f - Frequency - Hz

Figure 29

0.01
0.001

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY
. 10
VOO=3.3V
RL=80
SE -

Voo = 3.3 V
PO=100 mW
RL=80
SE

I
+

I

~

I

~

j

"'"

1111
20

100

I

II

1k
f - Frequency - Hz

Figure 31

/'

f-

"

10k 20k

~111~50~W

20

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

/'
./~

I I

Po=25mW

-

0.01

~TEXAS

3-290

I

I ffll

~

-

AV=-1 VN

JJ.

PO= 100 mW

0.1

AV =-5 VN

1m.
0.01

". /

. , / ...... ~

AV=-10VN
0.1

J

0.1

Figure 30

10

I

J

0.01

Po - Output Power - W

TOTAL HARMONIC DISTORTION PLUS NOISE

~

1'-00.
f = 100 Hz

Q

~

100

1k
f - Frequency - Hz

Figure 32

10k 20k

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TiPiCAL CHARACTERiSTiCS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

'it

'it

j

3l

10
Voo = 3.3 V
PO=30mW
RL=320
SE

I

I

'0
z
+

+

c

I
I

~

~

AV= 10VN

~0

I
I I Iill -l-

I I 11111

I!

j

0.1

,/

AV=-5VN

~
I

I

Z

Z

..:.: 100 Hz

i

AV= 1VN

a+
j!:

111m-I

0.01 L..-....L.....J....J"-I...LJ.J.LL_.J.-I....L..I...I..1-W---I--L..u...u.w
0.001
0.01
0.1
Po - Output Power - W

/
20

Figure 33

vs
OUTPUT POWER

j
c

i!

+

0

~I

~

0.1

Po=20mW
PO=30mW

*-

(Y

a+
:z:

i'" I

~

~

0.1

~
:z:

....!"'f
0.01

0

~

-I I

Z

...

~ ~ f=20kHz

0

0

I!

VOO=3.3V
RL=320
SE

I

i!

~

10

'it

+
c

.!o!
c

10 k 20 k

TOTAL HARMONIC DISTORTION PLUS NOISE

FREQUENCY
VOO=3.3V
RL=320
SE

j

1k
f - Frequency - Hz

vs
10

I

100

Figure 34

TOTAL HARMONIC DISTORTlON PLUS NOISE

'it

II

.1111

0.01

I!S5 ~f=1kHZ
~ 1"".,1.::, 20 Hz

!

~

0.01

~

I

PO=10mW

z+

a

...:z:

II IIII

0.001
20

100

1k

10k 20k

0.001
0.001

f - Frequency - Hz

Figure 35

0.1
0.01
Po - Output Power - W

Figure 36

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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

100

vs

FREQUENCY

FREQUENCY

100

~VDD=5V
f- BW = 22 Hz to 22 kHz
f- RL=4n

f>

Center

I

~

VDD = 3.3 V
BW = 22 Hz to 22 kHz
RL=4n

'iii

=-

t
I

OUTPUT NOISE VOLTAGE

vs

~
=-

r-

U"

10

Right

III

f

r-

Iz

r--

I
I

!

>

>

lull
10

I

c

1

1

20

100

10k 20k

1k

100

20

f - Frequency - Hz

Figure 38

SUPPLY RIPPLE REJECTION RATIO

0

I

I
c

vs

FREQUENCY

FREQUENCY

0

"'

-20

ID

-30

a:

I

~
D.
D.

Jl

-40
-60

./'
V

-«I
VDD=3.3V

-70

Ic

-30

11111ll/

-«I 100.

-100
20

100

I

-60

r'lljIlI

I'-.

-60

III

8:

Jl

V~D'~~~II

-80

J

-40

~

~

SE
-20

.2

RL=4n
Ca=4.7I1F

-10

"
I
0

0

IIi'

SUPPLY RIPPLE REJECTION RATIO

vs

RL=4n
CB = 4.7 I1F
BTL

-10

VDD=5V

"'"

N

-70

-«I
-90

IIIIII
1k

10 k 20k

"

./

VDD = 3.3 V

-100
20

f - Frequency - Hz

100

1k
f - Frequency - Hz

Figure 39

Figure 40

~TEXAS

3-292

10k 20k

1k
f - Frequency - Hz

Figure 37

"

I-I--

Right

'!i

c

ID

r-

Center

I

II

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75285

10k 20k

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYPiCAL CHARACTERiSTiCS

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY
-40

-40

VOO=5V
_ PO=75mW
-50
RL=32Q
SE
-60

m -70

-60

.....

m

'Q

I
(J

.....

-70

'Q

I
.II:

VOO=3.3V
Po=35mW
RL=32Q
SE

-50 -

I

..... ....

-60
..... r--

-90

.II:

Left to Right

'

.....

.......

-100

J

~

-120
20

.....

-90

,

Left to Right

t?-

>

......

Right to Left .....

"'

-110

......

-100

> =::;

Right to Left .....

roo..

-60

-110

111111
100

111111
100

-120

10 k 20 k

1k

20

f - Frequency - Hz

:::::~

1k

10 k 20 k

f - Frequency - Hz

Figure 41

Figure 42

OPEN LOOP RESPONSE
100
VOO=5V
BTL

180°

80
~

60

m

'Q

~

40

r--.

I

c

a;

CJ

11111111
Phase

I":

111

.....

CD

III
III

.c

GaU

20

90°

D-

0°

.....

0

.....
-90°

-20
-40
0.01

0.1

10

100

1000

-180°
10000

f - Frequency - kHz

Figure 43

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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOSl67A - .JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISnCS
OPEN LOOP RESPONSE

60

-

~

1SOO

IIIII Voo
BTL :3.3 V
111111

IIII~~·

All

"

III
'a

I

c

a

",v

rn
I~ .II

"

"

INIIiI
~

-",v

-40

0.01

0.1

10
100
f - Frequency - kHz

1000

-1SOO
10000

Figure 44
CLOSED LOOP RESPONSE
10
VDD=5V
AV=-2VN
PO=1.5W

9

_45°

BTL

8
7

_90°

Gain
III
'a
I

i

J

6

II

If'

5
4

Phase

3

-1SOO

If'

2

-225°

o

20

-2700
100

1k
10k
f - Frequency - Hz

100k 200k

Figure 45

~1ExAs

3-294

-135°

INSTRUMENTS
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j

a.

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULV 1997-REVISEDMARCH 2000

TYPiCAL CHARACTERiSTiCS
CLOSED LOOP RESPONSE
10

VoD'~'3.3V

9

AV =-2 VN
Po = 0.75W
BTL

8
7

I

Gain
ID

6

'a

I

c

~

1/

~

5

_90 0

-1350

j

A.

4

...

Phasa

_1800

3
~

2

o

20

,

-2700
100k 200k

10k
1k
f - Frequency - Hz

100

-2250

Figure 46
CLOSED LOOP RESPONSE
0

~JJI

II

-1

/

-2

00

_45 0

-3
ID

-900

-4

7

'a

I

-5

CJ

-e

c
'ii

-1350

J

Phase

-1800

-7

)~

-8

VOO=5V
AV=-1 VN
PO=0.5W
SE

-9

11111

-10
20

100

-2250

111111111

10k
f - Frequency - Hz
1k

_270 0
100k 200k

Figure 47

~TEXAS

INSTRUMENTS
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3-295

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997-REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
0
-1

I

-2

0°

b~l~ 1

~

_45°

If

-3
III
'1:1
I

c
iii
CJ

_90°

-4

I

-Q

J

-135°

-6

a.

Phase
_180°

-7

)~

-6

VDD=3.3V
AV=-1 VN
PO=0.25W
SE

-9

11111

-10

100

20

I 11111111

-

_225°

-270°

100k 200k

10 k
f - Frequency - Hz

1k

Figure 48
SUPPLY CURRENT

OUTPUT POWER

vs

vs

SUPPLY VOLTAGE

SUPPLY VOLTAGE
3

30

2.5

25

~

~
I

I

~

J

iOJ

'S

;-

THb+N = 110/0
BTL
Center Channel

I

I

Q

rP

3

______

~

________

~

______

4
5
VDD - Supply Voltage - V

~

6

/

/
o

2.5

/
".

/

V

,/

V

~RL=SO

"
3

3.5

4

4.5

5

VDD - Supply Voltage - V

Figure 49

Figure 50

~TEXAS

3-296

V

RL=40
1.5

0.5

O~

/"

2

0

E

.L.

INSTRUMENTS
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5.5

6

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYpiCAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

vs

vs

SUPPLY VOLTAGE

LOAD RESISTANCE
3

THO+N=1%
SE
Each LIR Channal

\

2.5

0.8
~
I

I

'5
So
~

0

RL=40/
0.6

,

0.4

~
./

,p

./
V V

0.2

~

.
I

'5
0

\.
'\

I

,p
0.5

....-

o

o

3

2.5

3.5
4
4.5
5
VOO - Supply Voltaga - V

1\\

1.5

t

V

,/

RL=320

\

2

I0

IL

RL=80

V . /' /

I

V

THO+N=1%
BTL
Centar Channel

\

5.5

6

o

4

I
~

0

vs

LOAD RESISTANCE

OUTPUT POWER

I

,p
0.2

o

~

1.2
~
I

c

I

0

\ \.

28

32

RL=40

/

--...

I ftL.oo -..........
I
0.6

\

VOO=5V

1'.., .....

,,~

VOO = 3.3 V

o

24

°

1.4

0.8

\

20

POWER DISSIPATION

0.6

0.4

16

r--

Figure 52

THO+N=1%
SE
Each LJR Channel

\

12

RL - Load Resistanca -

\\

'5

8

vs

\

I

...............

...........
VOO=3.3V

OUTPUT POWER

~

"~

'" --- --

Figure 51

0.8

,O=5V

----

I

~

r--

--

I"

I

4

8
12
16
20
24
RL - Load Reslstanca -

°

28

0.4
VOO=5V
BTL
cantar Channel

0.2

32

o

o

Figure 53

0.5
1.5
Po - Output Powar - W

2

Figure 54

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TPA010a
1.7S-W a-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION
va
OUTPUT POWER

POWER DISSIPATION
VB
OUTPUT POWER

0.8

~
I

0.6

0.6

RL=~

c

i

iii

~

0.4

I

Do

I
CI

Do

0.2

o

t

V

V

--

0.5

--

~
I

c

i

I

iii

is

--

RL=80

~

0.4

I
I

0.3

~
o

0.1

0.2

~

VOD = 3.3 V
BTL
Center Channel

SE

Each UR Channel

0.3

0.2

OA

0.5

o

0.6

w

_L

o

0.25

Figure 56
POWER DISSIPAnON
VB
OUTPUT POWER

0.6.------r--...,---..,-----,r----,
VDD=3.3V

--

SE

Each UR Channel

I

~

i ..

RL=40

I

I

0.5

Po - Output Power -

Figure 55

~

..........

I

0.1

VOD=5V

Po - Output Power -

0.21--~~.c:.--_+_--+----I--__I

~

0.05

0.1

0.15

0.2

Po - Output Power - W

Figure 57

~TEXAS
3-298

RL=40

I/RL=SO--"" 1'- .....

~
RL=320

/

-~~

v

INSTRUMENTS

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0.25

0.75

w

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 58)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Side View (e)

End View (b)

Bottom View (e)

Figure 58. Views of Thermally Enhanced PWP Package

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TPA0103
1.75-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
bridged-tied load versus single-ended mode
Figure 59 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA01 03 center -channel BTL
amplifier consists of two linear amplifiers driving both ends of the load. There are several potential benefits to
this differential drive configuration but initially consider power to the load. The differential drive to the speaker
means that as one side is slewing up the other side is slewing down and vice versa. This in effect doubles the
voltage swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power
equation, where voltage is squared, yields 4x the output power from the same supply rail and load impedance
(see equation 1).
V

_ VO(PP)
(rms) -

212

Power -

V(rms)

2

(1)

-At
VDD

J'

~

VO(Pp)

Figure 59. Bridge-Tied Load Configuration

In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-0 speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration of the UR channels as shown in Figure 60. A coupling capacitor
is required to block the dc offset voltage from reaching the load. These capacitors can be quite large
(approximately 33 I1F to 1000 I1F) so they tend to be expensive, heavy, occupy valuable PCB area, and have
the additional drawback of limiting low-frequency performance of the system. This frequency limiting effect is
due to the high pass filter network created with the speaker impedance and the coupling capacitance and is
calculated with equation 2.

~TEXAS

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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

APPLiCATiON iNFORMATiON
fc =

(2)

1
2nRL Cc

For example, a 68-I1F capacitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

Voo

~dB~----~~====
+- te =73 Hz, 32 0, 68 I1F

te

Figure 60. Single-Ended Configuration and Frequency Response

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be -calculated by subtracting the RMS value of the output voltage from Voo.
The intemal voltage drop multiplied by the RMS value of the supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in
the amplifier, the current and voltage waveform shapes must first be understood (see Figure 61).
100

,/

V(LRMS)

---fVt/V"Vffll-

IOO(RMS)

Figure 61. Voltage and Current Waveforms for BTL Amplifiers

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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most cif the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

PL
Efficiency = - P sup

(3)

Where:

= Voo loorms

Vpp

= :It RL
Efficiency of a BTE Configuration

P

V pp 2

P sup

2RL

L
= -= -- x

:It

RL

VooVpp

Vpp:lt

:ltJ2P R

2Voo

2Voo

L L
= - - = --'::-:-:--==--=

(4)

Equation 4 can also be used for SE operations.
Table 1 employs equation 4 to calculate efficiencies for four different output power levels. Note thatthe efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.

Table 1. Efficiency Vs Output Power in 5-V 8-0 BTL Systems
OUTPUT POWER
(W)

EFFICIENCY
(%)

PEAK-TO-PEAK
VOLTAGE

INTERNAL
DISSIPATION

(V)

(W)

0.55

44.4

2.00
2.83

62.6

4.00

0.59

70.2

4.47t

0.53

0.25
0.50

31.4

1.00
1.25

0.62

t High peak voltages cause the THO to Increase.
A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 4, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up. As the numerator values of RL and PL decrease, efficiency
decreases.

-!I1TEXAS
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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997 - REVISED MARCH 2000
•

__ I

1 - . . . . . .-... . . . . . . _ _ _ . . . . . . . .,... • •

" ...... L. ...." •• V ........... rvn ..."

•• v .....

For example, if the 5-V supply is replaced with a 3.3-V supply (TPA01 03 has a maximum recommended Voo
of 5.5 V) in the calculations of Table 1 then efficiency at 0.5 W would rise from 44% to 67% and internal power
dissipation would fall from 0.62 W to 0.25 W at 5 V. Then for a stereo 0.5-W system from a 3.3-V supply, the
maximum draw would only be 1.5 W as compared to 2.24 W from 5 V. In other words, use the efficiency analysis
to chose the correct supply voltage and speaker impedance for the application.

selection of components
Figure 62 and Figure 63 are a schematic diagrams of typical computer application circuits.

CFCt RFC

~
6

C<

< RILC
CBY

19
NC 9
8

CIN

BYPASS

SHUTDOWN

21

RHPIN
RLiNEIN

NC

-j

11

RIL

RFR

5
4

RFL

LHPlN
LLiNEIN

RM1
100 kll

VDD 718 V
DD
HP/LINE 16

~

J-ROUT 22

COUTR
If
1\

+

RM3
1 kll

~
Left
MUX

Interna
Speaker

VDD
RM2
VDD 100kll

T

CIR

I
I
CIL

I-......

MODE A 14

r--Right
MUX

. J""t'"

COUT- 15

~MODEB

.,rvvv

NC 20

-j IL

)f
I

MUTE OUT

-lRIR

r--

COUT+ 10

'-

""-

---- .t

~

LOUT 3

+

1,

-=-

I'

COUTL

GNDIHS
1, 12, 13, 24

Figure 62. TPA0103 Minimum Configuration Application Circuit

~TEXAS

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3-303

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOSl67A-JULY 1997 - REVISED MARCH 2000

APPLICATION INFORMATION

CF~t~

5pF

Mono

RIC
10kn

IL

II
CIC
0.1 j1f

RFC100kn

6
19

:::k

!

CB
4.7JiFT
718
_ VDD

CIN

1 r>t

BYPASS

AC97

MODEB
System { Active/Shutdown 11
Control High/Low Gain
16

RIRHP
10kn
Right
Line

20
21

CIR
0.1 JiF

RLiNEIN

Right
MUX

...........

40
Internal
Speaker

RM2
100kn
<_NoteA)

,A

~

ROUT 22

I
COUTR
470JiF

If
1\

'-

RM3
1 kn

-=

RFRHP
10kn.

Left
Line

CIL
0.1 JiF

RILL
10kn

5

LHPIN

-

4

LLiNEIN

Left
. MUX

-

-

+
GND/HS
1,12,13,24

LOUT 3

J.

RFLHP
10kn
RFLL
50kn

NOTE A. This connection is for ultralow current In shutdown mode.

Figure 63. TPA0103 Full Configuration Application Circuit

~TEXAS

INSTRUMENTS
3-304

-=
40-32 o
Speakersor
Headphonea

RFRL
50kn
RILHP
10kn

VDD

RM1
100kn

MUTE OUT 11
SHUTDOWN 8

...L
-

RIRL
10kn

MODE A 14

CNTL

HP/LiNE

RHPIN

COUT- 15

,AWl,.

VDD

rr

COUT+ 10

POST OFFICE BOX 655303 • DAUAS, TEXAS 76265

ItCOUTL
470JiF

TPA010a
1.7S-W a-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

AppliCATiON iNFORMATiON
gain setting resistors, RF and R,
The gain for each audio input of the TPA01 03 is set by resistors RF and RI according to equation 5 for BTL mode.
BTL Gain = -

2(~~)

(5)

In SE mode the gain is set by the RF and RI resistors and is shown in equation 6. Since the inverting amplifier
is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included.
SE Gain

= -

(~~)

(6)

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA0103 is a MOS amplifier, the input impedance is very high,
consequently input leakage currents are not generally a concern although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values are required for proper startup operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 kn and 20 kn. The effective impedance is calculated in equation 7.
Effective Impedance

=

R R
R :
F

~

(7)

I

As an example consider an input resistance of 10 kn and a feedback resistor of 50 kn. The BTL gain of the
amplifier would be -1 0 and the effective impedance at the inverting terminal would be 8.3 kil, which is well within
the recommended range.
For high performance applications metal film resistors are recommended because they tend to have lower noise
levels than carbon resistors. For values of RF above 50 kil the amplifier tends to become unstable due to a pole
formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kn. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 8.

~dBF=====~~-----(8)

fc(lowpass)

fe

For example, if RF is 100 kn and Cf is 5 pF then fc is 318 kHz, which is well outside of the audio range.

~TEXAS

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3-305

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997- REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the corner frequency
determined in equation 9.

fc(highpass) =

21t~ICI

(9)

The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 k.Q and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 10.

C =_1_
I

(10)

21tRl f C

In this example, CI is 0.40 ~F so one would likely choose a value in the range of 0.47 ~F to 1 ~F. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI' CI) and
the feedback resistor (RF) to the load. This leakage current creates a dc offset voHage at the input to the amplifier
that reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capaCitor is the best choice. When polarized capacitors are used, the positive side of the capaCitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. Please note that it is important to confirm the capacitor polarity in the application.

power supply decoupllng, Cs
The TPA01 03 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 ~F placed as close as possible to the device Voo lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 ~F or greater placed near
the audio power amplifier is recommended.

~TEXAS

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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997- REVISED MARCH 2000

APPLiCATiON iNFORiviATiON
mldrall bypass capacitor, CB
The mid rail bypass capacitor, Ce, serves several important functions. During startup or recovery from shutdown
mode, CB determines the rate at which the amplifier starts up. The second function is to reduce noise produced
by the power supply caused by coupling into the output drive signal. This noise is from the midrail generation
circuit internal to the amplifier. The capaCitor is fed from a 25-kn source inside the amplifier. To keep the start-up
pop as low as possible, the relationship shown in equation 11 should be maintained.
(11 )

As an example, consider a Circuit where Ce is 0.1 ~F, CI is 0.22 ~F and RI is 10 kn. Inserting these values into
the equation 10 we get 400 ~ 454 which satisfies the rule. Bypass capacitor, Ce, values of 0.1 ~F to 1 ~F ceramic
or tantalum low-ESR capacitors are recommended for the best THD and noise performance.

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 12.

(12)

fc(high)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency comer higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 ~F is chosen and loads vary from 4 n, 8 n,
32 n, to 47 kn. Table 2 summarizes the frequency response characteristics of each configuration.

~1ExAs

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3-307

TPA0103
1.75-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SlOS167A-JUlY 1997 -REVISED MARCH 2000

APPLICATION INFORMATION
output coupling capacitor, Cc (continued)
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode

Cc

LOWEST FREQUENCY

40

33011F

120Hz

80

330l1F

80Hz

320

330l1F

15Hz

47,0000

33OI1F

0.01 Hz

RL

As Table 2 indicates, most of the bass response is attenuated into a 4-0 load, an 8-0 load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.
The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the relationship shown in equation 13.
1
(C B x 25

kn)

s_1_~_1_

(CIR I)

RLCC

(13)

mode control resistor network, RM1, RM2, RM3
Using a readily available 1/8-in. (3.5-mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed, the 100-kn/1-kn divider (see Figure 64) pulls the MODE A input low. When a plug is
inserted, the 1-kn resistor is disconnected and the MODE A input is pulled high. When the input goes high, the
center BTL amplifier is shutdown causing the speaker to mute. The SE amplifiers then drive through the output
capacitors (Co) into the headphone jack.

Input MUX operation
The HPILINE MUX feature gives the audio designer the flexibility of a multichip design in a single IC (see
Figure 64). The primary function of the MUX is to allow different gain settings for different types of audio loads.
Speakers typically require approximately a factor of 10 more gain for similar volume listening levels as
compared to headphones. To achieve headphone and speaker listening parity, the resistor values would need
to be set as follOWS:
Gain(HP) = _ (RF(HP»)
RI(HP)

(14)

If, for example RI(HP) = 20 kn and RF(HP) = 20 kn then SE Gain(HP) =-1
G .
(RF(LlNE»)
aln(LlNE) = R1(LlNE)

(15)

If, for example RI(LlNE) = 10 kn and RF(LlNE) = 100 kn then Gain(LlNE) = -10

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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

AFFLiCAiiOi~

ii"FORiwiAiiOi,J

Input MUX operation (continued)

RFRHP

~E RIRLINE

r

RFRLlNE

21

RLINEIN

20

RHPIN

r--COUTR
MUX

~L

II

P

~r-

RIRHP

~'

ROUT 22

Right Channel

If
1\

MID

VDD

J,

<

"

MODE A 14

System
Control

16

~P/LINE

CNTL
MODES 11

VDD

iF

Left Channel

Figure 64. TPA0103 Example Input MUX Circuit
Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides
the optimum distortion performance into the headphones where clear sound is more important.

mute and shutdown modes
The TPA01 03 employs both a mute and a shutdown mode of operation designed to reduce supply current, 100,
to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input
terminal should be held low during normal operation when the amplifier is in use. Pulling SHUTDOWN high
causes the outputs to mute and the amplifier to enter a low-current state, 100 = 5 ~. SHUTDOWN should never
be left unconnected because amplifier operation would be unpredictable. Mute mode alone reduces 100 <1 mAo

~TEXAS

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3-309

TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

APPLICATION INFORMATION

mute and shutdown modes (continued)
Table 3. Shutdown and Mute Mode Functions
OUTPUT

INPUTSt

AMPUFIER STATE

MODE A

HPILINE

MODEB

SHUTDOWN

MUTE OUT

{NPUT

OUTPUT

Low
X
X
Low
High

Low
X
X
High
Low

Low
High
Low
Low

Low
High
Low
Low
Low

Low
High
High
Low
High

LJR Line
X
X
LJRHP
LJR Line

3 Channel
Mute
Mute
3 Channel
Mute

High

High

Low

Low

High

LJRHP

Mute

Low
Low
High
Low
High
High
High
Low
High
High
High
High
t Inputs should never be left unconnected.
X = do not care

Low
Low
Low

Low
Low
Low
Low

LJR Line
LJRHP
LJR Line
LJRHP

Center BTL
Center BTL
LJRSE
LJRSE

-

Low

using low-ESR capaCitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

5-V versus 3.3-V operation
The TPA01 03 operates over a supply range of 3 V to 5.5 V. This data sheet provides full specifications for S-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus S-V operation as far as supply bypassing, gain setting, or stability goes.
For 3.3-V operation, supply current is reduced from 19 mA (typical) to 13 mA (typical). The most important
consideration is that of output power. Each amplifier in TPA0103 can produce a maximum voltage swing of
VOO -1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) 2.3 V as opposed to VO(PP)
= 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into an 8-0 load before
distortion becomes significant.

=

Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power for a given output-power level than operation from S-V supplies.
When the application demands less than 500 mW, 3.3-V operation should be strongly considered, especially
in battery-powered applications.

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TPA0103
1.75-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A - JULY 1997 - REVISED MARCH 2000

APPliCATiON iNFORMATiON
headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA0103 data sheet, one can see that when the
TPA01 03 is operating from a 5-V supply into a 4-0 speaker that 2 W RMS levels are available. Converting watts
to dB:
P dB

10Log ( : : )
10Log

(~)

3 dB
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
3 dB - 15 dB = - 12 dB (15 dB headroom)
Converting dB back into watts:
1QPdB/10 x P
- 12 dB

=

ref

63 mW (15 dB headroom)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 1.5 W of continuous power output with 0 dB of
headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 5-V, 4-0 system, the internal dissipation in the TPA0103
and maximum ambient temperatures is shown in Table 4.

Table 4. TPA0103 Power Rating, 5-V, 4-0., Three Channel
CONFIGURATION
Center only, Po = 2 W max

LIR only. Po = 500 mW max
Center, Po = 2 W max
and
LJR • Po = 500 mW max

HEADROOMT

POWER DISSIPATION
2 x LlR + CENTER

=TOTAL

TA(MAX)*
35°CIW

25°CIW

OdB

0

1.25W

1.25W

BloC

93°C

15dB

0

0.6W

0.6W

104°C

110°C

OdB

0.6W

0

1.2W

83°C

95°C

15dB

0.2W

0

O.4W

111°C

115°C

OdB

0.6W

1.25W

2.45W

39°C

63°C

15dB

0.2W

0.6W

lW

90°C

100°C

t The 2 W max at 0 dB IS a maximum level tone that IS very loud. 15 dB IS a typical headroom requirement for musIc.
:I: This parameter is based on a maximum junction temperature (TJ) of 125°C.

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TPA0103
1.7S-W 3-CHANNEL STEREO AUDIO POWER AMPLIFIER
SLOS167A-JULY 1997- REVISED MARCH 2000

APPLICATION INFORMATION
headroom and thermal considerations (continued)
DISSIPATION RATING TABLE

=

=

PACKAGE
pwpt

TAS25°C
2.7W

DERATING FACTOR
21.8mWrC

TA 70°C
1.7W

TA 85°C
1.4W

pwp:j:

2.8W

22.1 mWrC

1.8W

1.4W

t This parameter Is measured with the recommended copper heat sink pattem on a l-Iayer PCB, 41n2 5-in x 5-ln PCB, 1 oz.
copper, 2-ln x 2-ln coverage.
:j: This parameter Is measured with the recommended copper heal sink pattem on an 8-layer PCB, 6.9 In2 1.5-ln x 2-in PCB,
1 oz. copper with layers 1, 2, 4, 5, 7, and 8 al 5% coverage (0.9 In2) and layers 3 and 6 all00%coverage (6 in2).

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 LFM
and 300 LFM data from the dissipation rating table, the derating factor for the PWP package with 6.9 in2 of
copper area on a multilayer PCB is 22.1 mW/oC and 53.7 mW/oC respectively. Converting this to 0JA:
Derating
For 0 LFM:

22.1 mW/oC
= 45°C/W
For 300 LFM:

53.7 mW;oC
= 18°C/W

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled forthe two SE channels and added to the center channel
dissipation. Given 0JA, the maximum allowable junction temperature, and the total internal dissipation, the
maximum ambient temperature can be calculated with the following equation. The maximum recommended
junction temperature for the TPA0103 is 150°C. The internal dissipation figures are taken from the Power
Dissipation vs Output Power graphs.
TA Max = T J Max - 9 JA Po
125 - 45(0.2 x 2

+ 0.6)

80°C (15 dB headroom, 0 LFM)

125 - 18(0.2 x 2

+ 0.6)

107°C (15 dB headroom, 300 LFM)

NOTE:
Internal diSSipation of 1 W is estimated for a 3-channel system with 15 dB headroom per channel
(see Table 4 for more information).

Table 4 shows that for most applications no airflow is required to keep junction temperatures in the specified
range. The TPA01 03 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. However, sustained operation above 125°C is not
recommended. Table 4 was calculated for maximum listening volume without distortion. When the output level
is reduced the numbers in the table change significantly. Also, using 8-n speakers dramatically increases the
thermal performance by increasing amplifier efficiency.

~TEXAS

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INSTRUMENTS
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TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997-

• High Power with PC Power supply
- 1.5W/Chat5V
- 600 mW/Ch at 3 V
• Ultra-Low Distortion
< 0.05% THD+N at 1.5 Wand 4-0 Load
• Bridge-Tied Load (BTL) or Single Ended
(SE) Modes
• Stereo Input MUX
• Surface-Mount Power Package
24-Pin TSSOP PowerPADTM
• Shutdown Control ••• IDD < 10 I1A

CFR

CIR

PACKAUi:

{TOP VIEW)

GNDIHS
NC
LOUT+
LLiNEIN
LHPIN
LBYPASS
LVOO
SHUTDOWN
MUTE OUT
LOUTMUTE IN
GNDIHS

10
2
3
4
5
6

24
23
22
21
20
19

7

18

8

17
16
15
14
13

9
10
11
12

NC

21

RUNEIN

20

RHPIN

ROUT+ 22
ROUT- 15

19 RBYPASS

car
-=

System
Control

CoUTR

Voo

-=
11
9

MUTE IN
MUTE OUT

8 SHUTDOWN

100 kil

Bias, Mute,
Shutdown,
andSE/BTL
MUXControl

VOO
6
NC

-1

RIL

GNDIHS
NC
ROUT+
RLiNEIN
RHPIN
RBYPASS
RVOO
NC
HPILINE
ROUTSElBTL
GNDIHS

RFR

RIR

-1

PWP

LBYPASS

5 LHPlN
4

LUNEIN

1 kil

-=

E

LOUT+ 3

Left
MUX

LOUT- 10

CiL
CFL

..

RFL

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

~ Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

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Copyright © 2000, Texas Instruments Incorporated

3-313

TPA0102
1.5-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

description
The TPA0102 is a stereo audio power amplifier in a 24-pin TSSOP thermal package capable of delivering
greater than 1.5 W of continuous RMS power per channel into 4-0 loads. This device functionality provides a
very efficient upgrade path from the TPA4860 and TPA4861 mono amplifiers where three separate devices are
required for stereo applications: two for speaker drive, plus a third for headphone drive. The TPA01 02 simplifies
design and frees up board space for other features. Full power distortion levels of less than 0.1 % THD+N from
a 5-V supply are typical. This provides significant improvement in fidelity for speech and music over the popular
TPA4860/61 series. Low-voltage applications are also well served by the TPA0102 providing 600-mW per
channel into 4-0 loads with a 3.3-V supply voltage.
Amplifier gain is externally configured by means of two resistors per input channel and does not require external
compensation for settings of 2 to 20 in BTL mode (1 to 10 in SE mode). An intemal input MUX allows two sets
of stereo inputs to the amplifier. In notebook applications, where internal speakers are driven as BTL and the
line (often headphone drive) outputs are required to be SE, the TPA01 02 automatically switches into SE mode
when the SE/BTL input is activated. Using the TPA01 02 to drive line outputs up to 500 mW/channel into external
40 loads is ideal for small non-powered external speakers in portable multimedia systems. The TPA01 02 also
features a shutdown function for power sensitive applications, holding the supply current below 5 J.lA. In
speakerphone or other monaural applications, the TPA01 02 is configured through the power supply terminals
to activate only half of the amplifier which reduces supply current by approximately one-half over stereo
applications.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TPA01 02 to operate at full power into 4-0 loads at ambient temperature of up to
55°C. Into 8-0 loads, the operating ambient temperature increases to 100°C.
AVAILABLE OPTIONS
PACKAGE
TA

TSSOP
(PWP)

4O"C to 85°C

TPA0102PWP

~1ExAs

3-314

INSTRUMENTS
POST OFFICE BOX 655303 • DAU.AS. TEXAS 75285

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

Terminai functions
TERMINAL
NAME

NO.

110

DESCRIPTION

GNDIHS

1.12.
13,24

HPILINE

16

LBYPASS

6

LHPIN

5

I

Left channel headphone input, selected when HPILINE terminal (16) is held high

LLiNE IN

4

I

Left channel line Input, selected when HPILINE terminal (16) Is held low

LOUT+

3

LOUT-

10

0
0

Left channel- output in BTL mode, high-impedance state in SE mode

LVDD
MUTE IN

7

I

Supply voltage input for left channel and for primary bias circuits

11

I

Mute all amplifiers, hold low for normal operation, hold high to mute

MUTE OUT

9

0

Follows MUTE IN terminal (11), provides buffered output

NC

Ground connection for circuitry, directly connected to thermal pad
I

Input MUX control input, hold high to select LJRHPIN (5, 20), hold low to select LJRLlNEIN (4, 21)
Tap to voltage divider for left channel intemal mid-supply bias

Left channel + output in BTL mode, + output In SE mode

No Intemal connection

2,17,23

RBYPASS

19

RHPIN

20

I

Right channel headphone input, selected when HPILINE terminal (16) is held high

RLINEIN

21

I

Right channel line input, selected when HP/LINE terminal (16) is held low

ROUT+

22

Right channel + output in BTL mode, + output in SE mode

ROUT-

15

0
0

RVDD
SElBTL

18

I

Supply voltage input for right channel

14

I

Hold low for BTL mode, hold high for SE mode

SHUTDOWN

8

I

Places entire IC in shutdown mode when held high, 100 < I !iA

Tap to voltage divider for right channel intemal mid-6upply bias

Right channel- output In BTL mode, high impedance state in SE mode

-!I TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

3-315

TPA0102
1.5-W STEREO AUDIO POWER AMPLIFIER
SLOSI66E - MARCH 1997 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... intemally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those Indicated under "recommended operating conditions· is not
Implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
PWP

DERATING FACTOR
2.7VV:1:

21.8mWI"C

1.7W

1.4W

:1: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMAOO2), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the Information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO

Operating free-air temperature, TA

Common mode input voltage, VICM

MIN

NOM

MAX

3

5

5.5

VOO=5V,
250 mW/ch average power,

4-0 stereo BTL drive,
With proper PCB design

-40

85

VOO=5V,
1.5 W/ch average power,

4-0 stereo BTL drive,
With proper PCB design

-40

55

UNIT
V

°C

VOO=5V

1.25

4.5

VOO=3.3V

1.25

2.7

V

dc electrical characteristics, TA = 25°C

VOO=5V
100

rvpt

MAX

Stereo BTL

19

25

StereoSE

9

15

rnA

Mono BTL

9

15

rnA

TEST CONDITIONS

PARAMETER

VOO=3.3V

Veo

Output offset voltage
(measured differentially)

VOO=5V

IOO(MUTE)

Supply current in mute mode

VOO=5V

3

10

rnA

13

20

rnA

StereoSE

3

10

rnA

Mono BTL

3

10

rnA

MonoSE

3

10

rnA

5

25

mV

15

J1A
J1A

Gain =2,

100 in shutdown
VOO=5V
IOOCSO)
NOTE 1: At 3 V < VOO < 5 V the dc output voltage is approximately Vool2.

~1ExAs

3-316

rnA

Stereo BTL

MonoSE

Supply current

UNIT

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

See Note 1

800
5

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E- MARCH 1997 - REVISED MARCH 2000

ac operating characteristics, VDD

=5 V, TA =25°C, RL =4 Q

PARAMETER

Po

TEST CONDITIONS

Output power (each channel) see Note 2

MIN

TYP

THD=0.2%,

BTL

1.25

THD=1%,

BTL

1.5

THD=0.2%,

SE

500

THD=1%,

SE

600

MAX

UNIT
W
mW

THD+N

Total hannonic distortion plus noise

Po =1W,

f = 20 to 20 kHz

200

m%

80M

Maximum output power bandwidth

G= 10,

THD<5%

>20

kHz

BTL

72°

Phase margin

Open Load

71°

SE

52°

Power supply ripple rejection

f= 1 kHz

75

f= 20 -20 kHz,

60

Mute attenuation

dB

85

dB

65

dB

LinelHP input separation

100

dB

BTL attenuation in SE mode

100

dB

2

MQ

Channel-to-channel output separation

ZI

Input impedance

Vn

Output noise voltage

1= 1 kHz

Signal-Io-noise ratio

Po =500mW,

BTL

95

dB

25

I1V(nns)

NOTE 2: Output power is measured at the output tanninals 01 the IC at 1 kHz.

ac operating characteristics, VDD

=3.3 V, TA =25°C, RL =4 Q

PARAMETER

TEST CONDITIONS
THD=0.2%

Po

Output power (each channel) see Note 2

THD+N

Total hannonic distortion plus noise

BOM

Maximum output power bandwidth

BTL

Power supply ripple rejection

MAX

UNIT

600

THD= 1%

BTL

750

SE

200

THD=1%,

SE

250

Po =600mW,

1 = 20 to 20 kHz

250

m%

G=10,

THD<5%

>20

kHz

mW

92°

Open Load

70°

SE

57°

1= 1 kHz

70

1= 20 -20 kHz

55

Mute attenuation

dB

85

dB

65

dB

LinelHP input separation

100

dB

BTL attenuation in SE mode

100

dB

2

MQ

Channel-to-channel output separation

ZI

Input impedance

Vn

Output noise voltage

Signal-ta-noise ratio

NOTE 2

TYP

THD = 0.2%,

BTL
Phase margin

MIN

1 = 1 kHz

Po= 500 mW,

BTL

95

dB

25

I1V(nns)

Output power is measured at the output tenninals 01 the IC at 1 kHz.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • OALLAS, TEXAS 75265

3-317

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

4.7f1F

II
.1. CB

SElBTL -+--,

-=-

HP/LINE

Figure 1. BTL Test Circuit

~
VDD
SElBTL

-=-

HPILINE

Figure 2. SE Test Circuit

3-318

-!11
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

RL=4Q,8Q,or320

1

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
4,5,7,8,11,12,14,15,17,18,20,
21,23,24,26,27,29,30,32,33

vs Frequency

THO+N Total harmonic distortion plus noise

3,6,9,10,13,16,19,22,25,28,
31,34

vs Output power
Output noise voltage

vs Frequency

35,36

Supply ripple rejection ratio

vs Frequency

37,38

Crosstalk

vs Frequency

39-40

Open loop response

vs Frequency

43,44

Closed loop response

vs Frequency

45-48

Supply current

vs Supply voltage

49

Po

Output power

vs Supply voltage
vs Load resistance

50,51
52,53

Po

Power dissipation

vs Output power

Vn

100

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY
10

10

fit.

fit.

:: VOO=5V
- f=1kHz
- BTL

I

I

I

I

I

I

RL=80

~
-t!0

f!
:!

~
I

0.1

AV=-10VN

..........

AV=-20VN

,
........

I

"

+
Q

f\~

!

li
Z

+
c
0

J

0.1

Iz

~

RL=40

f!

•

I

:/

c

~

-t!0

VOO=5V
PO=1.5W
RL=40
BTL

I

+

:I:

54-57

AV=...!J.VN -

Z

111111111-

.:!i

j!:

j!:
0.01

o

0.25 0.5 0.75

1

1.25 1.5 1.75

2

2.25 2.5

~
V

0.01
20

100

1k

II IIII

10 k 20k

f - Frequency - Hz

Po - Output Power - W

Figure 3

Figure 4

~TEXAS

INSTRUMENTS
POST OFFICE BOX 665303 • DALLAS, TEXAS 75265

3-319

TPA0102
1.5-W STEREO AUDIO POWER AMPLIFIER
SlOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

fI!

va

FREQUENCY

OUTPUT POWER

10

fI!

VOO=5V
RL=40
AV=-2VN
BTL

I

J
!
+
c

10
VOO=5V
RL=40
BTL

I

Iz

+

c

0

~

i!

j

j

Q

PO=1.5W

S

I•
:z:

TOTAL HARMONIC DISTORTION PLUS NOISE

va

PO=0.75W -

/A
~

0.1

IIf

f= 20 kHz

Q

S

c

~

0

I

0.1

f= 1 kHz

'§

Po = 0.25W

z+
j!:
100

20

1k
f - Frequency - Hz

f=20Hz

Q

:z:

IIIIIII

I-

1111111

0.01

I

Z

+

1111111

Q

~

0.01
0.01

10k 20k

0.1
Po - Output Power - W

FigureS

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

fI!

va

FREQUENCY

FREQUENCY

fI!

VOO=5V
RL=80
AV=-2VN
BTL

J!

!

10
VOO=5V
PO=1W
RL=80
BTL

I

Iz

+

+

c
oS!

c

~

1:

is
I

TOTAL HARMONIC DISTORTION PLUS NOISE

va
10

I

10

i
w~ I
.~

c
0

l"""-

0

Po = 0.5
0.1

B

t--PO=1W

I

Z

......

,:!i
j!:

I
I I

0.01
20

100

II

~

II

"
V jOI=

1k
f - Frequency - Hz

0.1

AV=-10VN

AV =-20

VNa
L

I'

B
I

i.iil~1

Z

-

Av=-2VN -

,:!i
j!:

10k 20k

III

0.01
20

Figure 7

100

1k
f - Frequency - Hz

FigureS

~TEXAS

INSTRUMENTS
POST OFFICE B9X 655303 • DAllAS, TEXAS 75265

II Ull

10k 20 k

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

'#

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

10

~

VOO=5V
RL=80
AV =-2 VN
BTL

I

I
+

I

+

I

I

i

f=2OkHz

~
:c

0.1

0

II

0.01
0.01

~

I

0
j!:

~~~HzI

0.01

0.1
Po - OUtput Power - W

o

10

0.1

0.2 0.3 OA 0.5 0.6 0.7 0.8 0.9
Po - OUtput Power - W

10

VOO=3.3V
RL=40
AV =-2 VN
BTL

iz

+

c

c

0

~

;:

~=

V

l--' i-'"

l
0.1

I

AV =-20 VN ....

~

0

.r

~0
I§

Po = 0.7:;r: ~
0.1

Po = 0.1 W

i

~

I

0

LlJII

lill

0.01

100

Z

j!:

I

0.01
10k 20k

-

I II II

0

I II

1k

Po = 0.35W

I

Av=-2VN

20

i

::!

AV =-10 VN

Z

i!:

10

I

+

i
~
I
I

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
~

VOO =3.3 V
Po = 0.75W
RL=40
BTL

z

1

Figure 10

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

.I0

I

I

Figure 9

I

II

Z

N

~

0.1

1= ==

II

RL=80

I

f=1kHz

I

I
RL=40

~0

.....

Z

~

,

f- BTL

z

;:

I

~ VOo=3.3V

1= f=1kHz

.I0

(5

i
i
I

10

I

20

100

1k

f - Frequency - Hz

f - Frequency - Hz

Figure 11

Figure 12

II II
10 k 20k

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-321

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

il-

vs

OUTPUT POWER

FREQUENCY

10

+
c

1"""'--0

0

i
:z:

I

i'~

10
VOO=3.3V
PO=0.4W
RL=8n
BTL

I

j
z0

+

c

f= 20 kHz

i

OJ

is

~

~
as

il-

Voo= 3.3 V
RL=4n
AV =-2 VN
BTL

I

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

0.1

~0

....... i'~

~

f=1 kHz

:!
J

0.1

{].

I

+
Q

1111

j!:

r-

Z

I-toi.

j 111111

0.01
10

20

vs

il-

VOO=3.3V
I- RL=8n
I- AV=-2VN
f- BTL

~

+
c

V

19'

I

I

~

6

PO=0.4W

I

Z

j!:

Po = 0.1 W
0.01

20

100

1k

f=2OkHz

~0

:!

PO-O.25W

t--..

i

~

i-'

j!:

VOO = 3.3 V
RL=8n
AV =-2 VN
BTL

II

.!!

0

I

10

I

i!

0.1

10k 20k

TOTAL HARMONIC DISTORTION PLUS NOISE
OUTPUT POWER

I
J

1k

FREQUENCY

1=

+

100

vs
10

I

Will 11
Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

I

L

f - Frequency - Hz

Figure 13

il-

""

AV =-2 VN

:z:
r-

0.1
Po - Output Power - W

",

AV=-10VN

6

I

IIII

0.01
0.01

r-

I

f=2OHz

Z

~

AV=-2OVN

i','

10k 20k

0.1 I"---.

0.01
0.01

f - Frequency - Hz

Figure 15

f= 1 kHz
Ul

f~'20HZ

11111

I

0.1
Po - output Power - W

Figure 16

~1EXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

'i/.

10

vs

FREQUENCY

FREQUENCY

F

VOO=5V
PO=0.5W
f- RL=40
f- SE

'i/.

~

I

Iz

~
+

c
0

t-

0

1=
t- \'

'E

'&i

-

is
.!!

c

"""-

0.1

A

AV= 10VN

"/

/

I I ""
AV =-5 VN

~u

1111111

I

PO=0.5W

·2

I

0

~
::I:
S

~

Po =0.25 W

0.1

I

11~~1=-1 ~~

+
Q

::I:

0.01
20

100

I......

Z

+

Q

...
::I:

I IIII
1k
f - Frequency - Hz

~OI=I~·~I:

0.01
20

10k 20k

10

Iz

+

1=

~

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

VOO=5V
RL=40
AV=-2VN
SE

~

'i/.

tt-

~

Z

VOO=5V
PO=O.25W
RL=SO
SE

+

c

~

-0.1

1=

1=

II)

.s

.!!

f=2OkHz

S

10

I

~0

~
::I:

Q

.!!

c

j....

AV =-10 VN

0

~
::I:
iii

f =100 Hz

0.1

-I"I\.

I II III
AV=-5VN

~I

~I

Z

Z

...

...

::I:

f= 1 kHz

::I:

0.01
20

0.01
0.1
Po - Output Power - W

1"'""./

-

1/

AV= 1VN

~

+
Q
0.01
0.001

10 k 20 k

Figure 18

I"- .....

c

1k
f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

I

L

r--.
r--t-

100

Figure 17

'i/.

L.
~

I

~

I

Z

...

VOO=5V
RL=40
AV= 2VN
t- SE

Z

~0

S

t=
tt-

II)

c

::I:

10

I

+

~

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

100

lllll

1k

1

10k 20 k

f - Frequency - Hz

Figure 19

Figure 20

~TEXAS

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3-323

TPA0102
1.5-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

va

FREQUENCY

OUTPUT POWER

"#.

10

"#.

I

I

r= VOO=5V

I- RL=80

:=

J

Iz

t

8

+

r-

+

AV=-2VN
SE

'E

~

is

.....f=2OkHz

is

.!:!

~

I
,

TOTAL HARMONIC DISTORTION PLUS NOISE

va

~

~

III

~ Po=O.25W
~ 0.1~mmlm
~

0.1 W

I

=
II ~
'tI}o:-,....,-11 ......;~-_....u..L.U.L~
Po =0.05 W

~

"Po

j;
0.01

.......

I

~

Z

0.1

:E:

Z

~

j;

f=1ooHz
0.01
0.001

L....I..J...u.J.JUJ.---L-'-..........

20

100

1k

1=1 kHz

...

10 k 20 k

f - Frequency - Hz

l"'"'Nl
.J,.of
T -, II
Figure 22

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

10

I

~

Z

va

va

FREQUENCY

FREQUENCY

;:: VOO=5V
f: Po = 0.075 W
c- RL=320
r- SE

"#.

VOO=5V
RL=320
SE

:

"0
z
+
c
0

~0

'E

I

ic

co
·S

Ii

10

I

+
c

I

AV =-10 VN

./

AV =-6 VN

0.1

I

I ~~1~-1

j;

20

100

i
~

=

I

~NI

A

- -- -_._.

0.1

Z

Po=50mW
Po =75 mW
~

"V"rll

~

j;

1I1111

0.01

~0

:E:

~

Z
+
C

.-'PO=25mW
0.01

1k
f - Frequency - Hz

10k 20k

20

100

1k
f - Frequency - Hz

Figure 24

Figure 23

~TEXAS

INSTRUMENTS

3-324

0.1

0.01
Po - OUtput Power - W '

Figure 21

"#.

-'
'/

POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

10k 20k

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY
10

"'

'"
...

VDD=3.3V
PO=0.2W
RL=40
SE
~

.....
~.L....LJ""

-

Ilw.u-- i"""

0.1~~m11

0.1

II~YI~-1 r~

i"

~111""1

0.01 l..-..Lil:tl:l:tit:::::..L....L.W.lill_Ll...l..l.1JlU
0.001
0.01
0.1
Po - OUtput Power - W

111111

0.01

Figure 26

vs

FREQUENCY

OUTPUT POWER

=
:

VDD=3.3V
RL=40
SE

I

.1
~

-

~E

VDD=3.3V
RL=40
AV =-2 VN
SE

•
a
z

r--...

....

0

PO=0.2W

~0

t:0
//

Po =0.1 W

V
l/~

0.1

~

f= 20 kHz

~0

~
:%:

~

....

0.1

II
f=1kHz

i

~I

{!.

Z

+
Q

...

10

I

c

~

i

'#.

+

+
c

~
:%:

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10

t:0

10k 20k

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

0

1k

100

20

Figure 25

'#.

V

AV =-5 VN

I::::--+--HH-t+. f= 20 Hz /.-ftlHttt---t-++++l+H
t--...
I
I.JII f= 1 kHz -H-++I+H

.... ~

/

AV =-10 VN

f=2OkHz

:%:

0.01
20

--

100

I

~

Z

Po= 0.05 W

5"""

1111

1k
f - Frequency - Hz

+
Q

...

:%:

II
10k 20 k

---

f = 100 Hz
0.01
0.001

0.01

0.1

Po - Outpul Power - W

Figure 27

Figure 28

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3-325

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10

~

~

YOO=3.3Y
PO=100mW
RL=Sn
SE

I

J
+

10
YOO=3.3Y
RL=sn
SE

I

.~

Z

+
C

I

~
V

.2

J

V t:::~

Ay=-10YN
0.1

'li

/'

~

~

0

I-

is
.2
c

i

:c

!

~I

-

~

Ay=-5YN

I

11tH-

j!:

I

IIII

0.01
20

Ay=-1 YN

,;..

100

r--

"7

0.1

~

~I

I~II

PO=25mW

+

I-

0.01
20

-

Figure 30
TOTAL HARMONIC DISTORTION PLUS NOISE

+

~

vs

OUTPUT POWER

FREQUENCY
~

II

10
YOO=3.3Y
PO=30mW
RL=32n
SE

I

.!z

SE

f'-.. .....

C

vs

r==
YOO = 3.3 Y
r- RL=Sn
rr-

~

Z

I

+
c

II f= 1
20 kHz

~

Ic

Ic

Ay= 10YN

.2
c

~0

E

!

0.1

r-..... i'

!as

:c

f= 1 kHz

IIf

'li

~I

Z
+
C

:c

I-

0.01
0.001

I 111111

I IIIII....J0.1

z

Hz
t--! = 100
'"
I 1TIr-r

Ay= 1YN

c+
j!:

IIIII

V
0.01
20

0.01

0.1
Po - Output Power - W

,/

Ay=-5YN

100

1k
f - Frequency - Hz

Figure 31

Figure 32

~TEXAS

3-326

10k 20k

1k

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

10

~

100

Figure 29

~
I

~

I I

Z

C

10k 20k

1k
f - Frequency - Hz

r7

T~III~50~W

:c

II

II

Po= 100mW

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

II
10 k 20 k

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10

'#

.;0

••+
Z

+
C

0

i:
~

P'
0.1

{:.

i

-I

:z:

......
0.01

I

Z

~

....!"'I

'"I

:z:

~-f;'1 kHz

§

~ ....f.,:20Hz

i

0.01

{:.
I

PO=10mW

z+

~

i!:

0.1

0

Po=30mW

E
II

~

2ii
.2
c

Po=20mW

0

i

~ ~ f=20kHz

C

~
.~

VOO=3.3V
RL=320
SE

I

Z

i

10

'#

VOO=3.3V
RL=320
SE

I

Q

:z:

11 JlIl

0.001
20

~

100

0.001
0.001

10 k 20k

1k
f - Frequency - Hz

0.01
0.1
Po - Output Power - W

Figure 33

Figure 34
OUTPUT NOISE VOLTAGE

OUTPUT NOISE VOLTAGE

vs

vs

FREQUENCY

FREQUENCY

100

100
VOO=5V
BW = 22 Hz to 22 kHz
RL=4Q

I
:I.
I

E

-

"

10

Vo-

J

~
:I.
I

VOBTL

II
CI

VO+

~

I

'ii'
VOBTL

t

VOO = 3.3 V
BW = 22 Hz to 22 kHz
RL=4Q

Vo+

1l!
~

II

~

.~

-

-

II

10

Vo-

z

=
f

;:::=

0

I

I

C

c

>

>
1

1

20

100

1k
f - Frequency - Hz

10k 20 k

20

100

1k

10k 20k

f - Frequency - Hz

Figure 35

Figure 36

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-327

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY

SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY

0
III
'a
I

i

-20

c

-30

13

•
l

-40

ii!

-60

0

8:•

i

0
RL=40
CB=4.7I1F
BTL

-10

-10

.2

-20

'a
I

ic

-50

L

V"

Voo = 3.3 V
-70

IlllIL
...
!!II

:::I

III

III

-80 "-

...... i'"

I•

-90
100

f"'II~

"- ..... ""

-50

i

8:

-70

r--..l JJ

:::I

III

-80
-90

1k

-100

10k 20k

20

100

Figure 37

1k
f - Frequency - Hz

-60
III

'a

-70

I

i
0

-80

"r-..

-90

CROSSTALK
vs
FREQUENCY
-40

VOO=5V
PO=1.5W
RL=40
BTL

r-..

-50 t-60

IIILeft 10 Right

L

~
I'-.

i"

10k 20k

Figure 38

CROSSTALK
vs
FREQUENCY

-50 t-

L

Voo = 3.3 V

f - Frequency - Hz

-40

Voo=5V

>-

II 1111
20

SE

-40

-60

VOO=5V

-100

-30

RL=40
CB =4.7I1F

V

~

~
j.o'

RlghttoLeH

III

'a

I'~

-70

I

1
S

-100

VOO = 3.3 V
Po = 0.75W
RL=4Q
BTL

"

-80

r---1L~~
"'"

-90 t- Right to Left

L

V

""

V

V

I-"

-100

-110

-110

...
-120
20

100

1k
f - Frequency -

10k 20k

-120
20

Hz

1k
f - Frequency - Hz

Figure 39

Figure 40

~1ExAs
3-328

100

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS, lEXAS 75265

10k 20k

TPA0102
1.5-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CROSSTALK

CROSSTALK

va

va

FREQUENCY

FREQUENCY

-40

-40

VOO=5V
_ PO=75mW
-60
RL=32G

VOO=3.3V
Po=35mW
RL=32G

-50 -

SE

SE

-eO

-60
III

-70

'U

.....

1e
0

-60

-90

III

,

I

~

Left to RIght

III

-120
20

-60

0

-90

......
.......

,

Left to RIght

tt?-

-100

> :::::;

RIght to Left .....

-110

j

e

'?-

"

-100

.....

I

'r-..

..... 1'-.

-70

'U

RIght to

>-

Le~"""

-110

111111
100

1111111
100

-120
20

10k 20 k

1k

~;;

1k

f - Frequency - Hz

f - Frequency - Hz

Figure 41

Figure 42

10 k 20 k

OPEN LOOP RESPONSE

100

VOO~'5V
BTL

80

60
III

'U

~

IIIIIIII

...;:

Phase

II

40

"-

I

c

~

180°

......

20

90°

~

::

ll

III

.c

oaln

D..

0°

r--..

0

·1\11111

I'

_90°

-20
-40

0.01

0.1

10

100

1000

_180°
10000

f - Frequency - kHz

Figure 43

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-329

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOSl66E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN LOOP RESPONSE

-

80

60

11111

111111

?

Phase

c
iii

1\

goo

r..

I

11111 IT

40
III
'a
I

180°

YOo~'
3,3 Y
BTL

I~alll

20

1\

CJ

11\11111

0

~
-20
-40

0.01

0.1

10

100

1000

_180°
10000

f - Frequency - kHz

Figure 44
CLOSED LOOP RESPONSE

0°

10
YOO=5Y
Ay =-2 YN
PO=1.5W
BTL

9

8
7

_90°

Gain
III
'a
I

c

~

I

6

II

if'

5

Phase

-180°

3

i-"

2

-225°

o

20

100

1k

10k

_270°
100k 200k

f - Frequency - Hz

Figure 45

~TEXAS

3-330

-135°

J

II.

4

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
10
Voo = 3.3 V
Ay=-2VN
Po = 0.75W
BTL

9

8
7

L

Gain
III

6

I

5

'0

c

~

V

~

4

Phase

,

3
2

o

20

,
100

1k
10k
f - Frequency - Hz

-270"

100k 200k

Figure 46
CLOSED LOOP RESPONSE
0

/

-1

IL

-2

~a~~ I

-3
III

-4

V

'0

I

c

-6

CJ

-6

'ij

Phase

-7
VOO=5V
AV=-1 VN
PO=0.5W

/'

-8
-9

SE

1111

-10
20

100

I 11111111

1k
10k
f - Frequency - Hz

-270·
100k 200k

Figure 47

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-331

TPA0102
1.S-W STEREO AUDIO POWER AMPLIFIER
SLOS166E - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE

0

b~l~ I

/

-1

L

-2

-3
III
"0
I

_90°

-4

c

-0

CJ

-6

/
-135°

'iii

:I
.l!!

II.

Phase

'"

-7

VDD=3.3V
AV=-l VN
Po = 0.25W
SE

"

-8

--9

1111

-10
20

100

-

I 11111111

-270°
lOOk 200k

10k
1k
f - Frequency - Hz

Figure 48
SUPPLY CURRENT

OUTPUT POWER

vs

vs

SUPPLY VOLTAGE

SUPPLY VOLTAGE
3

30

~D+N=ll%

BTL
2.5 f- Each Channel

25

/V

JA:
E>

JA

=

1
Derating

=

0.J22

= 45°CjW
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given E>JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA01 02 is
150°C. The internal dissipation figures are taken from the Power DisSipation vs Output Power graphs.
TA Max

=
=

T J Max - E>JA Po
150 - 45(0.4 x 2)

=

114°C (15 dB headroom, 0 CFM)
NOTE:

Intemal disSipation of 0.4 W is estimated for a 1.5-W system with 15 dB headroom per channel.

Table 4 shows that for most applications no airflow is required to keep junction temperatures in the specified
range. The TPA01 02 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150°C to prevent damage to the IC. Table 4 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-0
speakers dramatically increases the thermal performance by increasing amplifier efficiency.

~TEXAS

INSTRUMENTS
3-348

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
• Compaiioie With PC 99 uesktop Line-Oui
Into 10-kO Load
• Internal Gain Control, Which Eliminates
External Gain-SeHing Resistors
• 2-W/Ch Output Power Into 3-0 Load
• PC-Beep Input
• Depop Circuitry
• Stereo Input MUX
• Fully Differential Input
• Low Supply Current and Shutdown Current
• Surface-Mount Power Packaging
24-Pin TSSOP PowerPADTM

P'NP PACKAaE
(TOP VIEW)

GND
GAl NO
GAIN1
LOUT+
LLiNEIN
LHPIN
PVOO
RIN
LOUTLIN
BYPASS
GND

10
2
3
4
5
6
7
8

9
10
11
12

24
23
22
21
20
19
18
17
16
15
14
13

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
VOO
PVOO
PCB ENABLE
ROUTSElBTL
PC-BEEP
GND

description
The TPA0112 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-0 loads. This device minimizes the number of
external components needed, simplifying the deSign, and freeing up board space for other features. When
driving 1 W into 8-0 speakers, the TPA0112 has less than 0.8% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
Amplifier gain is internally configured and controlled by way of two terminals (GAl NO and GAIN1). BTL gain
settings of -2, -6, -12, and -24 VN are provided, while SE gain is always configured as -1 VN for headphone
drive. An internal input MUX allows two sets of stereo inputs to the amplifier. In notebook applications, where
internal speakers are driven as BTL and the line outputs (often headphone drive) are required to be SE, the
TPA0112 automatically switches into SE mode when the SElBTL input is activated, and this reduces the gain
to -1 VN.
The TPA0112 consumes only 6 mA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to less than 150 /lA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TPA0112 to operate at full power into 8-0 loads at an ambient temperature of 85°C.

A.

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3-349

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B- MAY 1999- REVISED MARCH 2000

functional block diagram
RHPIN----t
RLiNEIN - - - - I

>--+-------

ROUT+

>--+--r-----

ROUT-

RIN - - - - - - - . , . . . - - - - \ - .

PC-BEEP
PC ENABLE

I---1

pc.
Beep

Power
Management

PVDD
VDD
BYPASS
SHUTDOWN

GAINO

GAIN1
SElBTL

' - - - - - GND

LHPIN---I
LLiNEIN - - - - I

>-.....- + - - - - -

LOUT+

>---+------

LOUT-

LIN - - - - - - - - - - - \ -..

~TEXAS

INSTRUMENTS
3-350

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

AVAILABLE OPTiONS

PACKAGED DEVICE
TA

TSSOP't
(PWP)

-40°C to 85°C

TPA0112PWP

t The PWP package is available taped and
reeled. To order a taped and reeled part, add
the suffix R to the part number (e.g.,
TPA0112PWPR).

Terminal Functions
TERMINAL
NAME
NO.

I/O

DESCRIPTION

BYPASS

11

GAINO

2

I

Bit 0 of gain control

GAIN1

3

I

Bit 1 of galn control

GNO

Tap to voltage divider for internal mid-supply bias generator

1,12,
13,24

Ground connectior for circuitry. Connected to the thermal pad.

LHPIN

6

I

Left channel headphone Input, selected when SElBTL is held high

LIN

10

I

Common left input for fully differential input. AC ground for single-ended inputs.

LLINEIN

5

I

Left channel line input, selected when SElBTL is held low

LOUT+

4

Left channel positive output in BTL mode and positive output In SE mode

LOUT-

9

0
0

Left channel negative output in BTL mode and high-impedance in SE mode

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > 1-V (peak-ta-peak) square wave is input
to PC-BEEP or PCB ENABLE is high.

PCB ENABLE

17

I

" this terminal is high, the detection circuitry for PC-BEEP is overridden and passes PC-BEEP through
the amplifier, regardless of its amplitude. " PCB ENABLE is floating or low, the amplifier continues to
operate normally.

7,18

I

Power supply for output stage

20

I

Right channel headphone input, selected when SElBTL is held high

RIN

8

I

Common right Input for fully differential input. AC ground for single-ended Inputs.

RLiNEIN

23

I

Right channel line input, selected when SElBTL is held low

ROUT+

21

Right channel positive output in BTL mode and positive output In SE mode

ROUT-

18

0
0

SHUTDOWN

22

I

Places entire IC in shutdown mode when held low, except PC-BEEP remains active

PVOO
RHPIN

Right channel negative output in BTL mode and high-impedance in SE mode

SElBTL

15

I

Input MUX control input. When this terminal Is held high, the LHPIN or RHPIN and SE output is selected.
When this terminal is held low, the LLINEIN or RLiNEIN and BTL output are selected.

VOO

19

I

Analog VOO input supply. This terminal needs to be isolated from PVOO to achieve highest perlormance.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-351

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS

SLOS204B - MAY 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted}t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, T J .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t Stresses beyond those listed under "absolute maximum ratings· may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions· is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
DERATING FACTOR

PACKAGE
PWP

2.7wt

1.7W

21.8mWI"C

l.4W

:j: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

(literature number SLMA002), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
High-level input voHage, VIH

MIN

MAX

4.5

5.5

SElBTL

4

SHUTDOWN

2

SElBTL

Low-level input voltage, VIL

3

Operating free-air temperature, TA

-40

v
V

0.8

SHUTDOWN

UNIT

85

v
°C

electrical characteristics at specified free-air temperature, Voo = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

VI=O, AV=2

Power supply rejection ratio

VOO=4 Vt05 V

IIIHI

High-level input current

VOO=5.5V,
VI=VOO

900

nA

IIILI

Low-level Input current

VOO=5.5V,
VI=OV

900

nA

100

Supply current

IVool
PSRR

BTL mode

6

8

3

4

150

300

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

mV
dB

77

SEmode

IODISO) Supply current, shutdown mode

3-352

25

UNIT

Output offset voltage (measured differentially)

mA

!LA

TPA0112
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS2048 - MAY 1999 - REVISED MARCH 2000

operating characteristics,

Voo =5 V, TA =25°C, RL =8 n, Gain =-2 ViV, BTL mode

PARAMETER

TEST CONDITIONS

THO = 1%,
RL=4n

f= 1 kHz,
f=20Hzt015kHz

Po

Output power

THO+N

Total harmonic distortion plus noise

PO=1W,

BOM

Maximum output power bandwidth

THO=5%

Supply ripple rejection ratio

f= 1 kHz,
CB = 0.4711F

SNR

I

BTL mode

Signal-to-noise ratio

Vn

Noise output voltage

ZI

Input Impedance

CB = 0.4711F,
f = 20 Hz to 20 kHz

I BTL mode

I SE mode

MIN

TYP

MAX

UNIT

1.9

W

0.75%
>15

kHz

68

dB

105

dB

16
30

I1V RMS

See Table 1

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE

vs Output power

1,4-7, 10-13,
16-19,21

vs Frequency

2,3,8,9,14,
15,20,22

THO+N

Total harmonic distortion plus noise

Vn

Output nO,ise voltage

vs Bandwidth

24

Supply ripple rejection ratio

vs Frequency

25,26

Crosstalk

vs Frequency

27-29

Shutdown attenuation

vs Frequency

30

Signal-to-noise ratio

vs Frequency

vs Output voltage

SNR

Closed loop respone
Po
Po

Output power
Power dissipation

23

31
32-35

vs Load resistance

36,37

vs Output power

38,39

vs Ambient temperature

40

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-353

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVIseD MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

10%
AV=-2VN
f=1 kHz
BTL
1I
I
il

••c+
Z

~

1%

i
15
I

I'

I

RL=40/

-=

RL=SO

.2

,

I

I

J

~
0.1%

II
RL=30

=-==

-

/

I

.P

I

z

+

~

Q

~
0.01%
0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 2.75

II IIII

0.01%
20

3

100

Po - Output Power - W

Figure 1

10%

RL=30
AV=-2VN
BTL

::
'0
z

+

~

:,;r-

--

%

Po =1.0W
0.1

If

0.1%

I

i'-ol"-

n

lllll
10k 20k

~
RL=30
AV =-2 VN
BTL

~

1k
f - Frequency - Hz

1

f=2OHz

eli

.\

100

........ f= 15 kHz

r-.-f=1 kHz

z

Po = 1.75W
0.01 %
20

i
I

1%

J

......

~

'E!

0

1M V

V

Po =0.5W

:-

+

c

0

.~

'.J'

II.

10k 20k

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

10%

I
I

1k
f - Frequency - Hz

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

I

AV=~V~

0.01%
0.01

0.1
Po - Output Power - W

Figure 4

Figure 3

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S204B - MAY 1999 - REVISED MARCH 2000

TYPiCAL CHARACTERiSTiCS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10%

10%

CD

f= 15 kHz

.!!!

z0

'"
I'-!!..= 15 kHz J

+

c

~

Ir

1%

i

rN

I III'

.J
1%

I

fl=~ kW

i"-

f = 1 kHz

.Il
c

......

,....."

0

t........

E
1\1

:z:

S

II

I........ r--

I, I II

~~OHZ

f=20Hz

0.1%

{;.

r-'
0.1%

I

Z

+

Q

I- RL=30
I- AV=-4iVN

~

0.01%
0.01

t-- RL=30
t-- Ay=-12VN

BTL

0.01%
0.01

0.1
Po - Output Power - W

BTL
0.1
Po - Output Power - W

Figure 6

Figure 5
TOTAL HARMONIC DISTORTION PLUS NOISE

~
z

+

c

~

..

.s

1%

vs

OUTPUT POWER

FREQUENCY
10%

f = 15 kHz

-

t-

t"--.."

f = 1 kHz

I

j

rr-

.Il

i
:z:

t"'-...

~

f=20 Hz
1......

1"-

0.1%

J

:!!

0.1

7

I

Z

z

I

~

10

0.01 %
20

Figure 7

V

r"\

AV=-2VN

V

,/

~

0.1
Po - Output Power - W

/

\ II
[.../ ~II

~

- RL=30
- Ay=-24YN
BTL

0.01%
0.01

Av= 12VN

,,,,r--.

S

{!.

AV =-24 YN /,,,,,,

%

Q

0

~

~

+

.Il
c

+
Q

PO=1.75W
RL=30
BTL

J

...,I

is

S

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10%

10

iVI=liI~1
100

1k
f - Frequency - Hz

10k 20k

FigureS

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-355

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE, GAIN SETTINGS
SLOS204B- MAY 1999- REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
va
FREQUENCY

TOTAL H~RMONIC DISTORTION PLUS NOISE
va
OUTPUT POWER

10%

10%

RL=4n
AV=-2VN
BTL

••+
Z

II

r-

C

~0

~

PO=1.5W

PO=1.0 Wf:::=

IJ
~

I

f=1 kHz

It+

0.1%

I

.1

f=2OHz

Z

L. ~

f=15kHz

i""oo

0

~~

Po = 0.25 W

r-.

1%

.2
c

~

"

RL=4n
AV=-2VN
BTL

+

Q

i!:
0.01%
20

100

1k

0.01%
0.01

10k 20k

0.1
Po - Output Power - W

f - Frequency - Hz

Figure 10

Figure 9
TOTAL HARMONIC DISTORTION PLUS NOISE
va
OUTPUT POWER

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

10%

10%

Iz

II

+

...........

c

~

"'
f=15kHz

~0

--....

~

~

t"oo..

:!
J
{!.

i

i
j

f=1 kHz

I

!

f=2OHz

+
Q

z

RL=4n
Av=-8VN
~ BTL

I-

i!:

0.01%
0.01

1% rf=1 kHz

.......
1""
0.1%

~

I I IIIII
0.1
Po - Output Power - W

10

RL=4n
AV=-12VN
BTL
~~ 1111111
0.01%
0.01
0.1
Po - Output Power - W

Figure 12

Figure 11

~TEXAS

3-356

r.......

..,

I II I
f~ml~

If

I

Z

....,I

.~

TI"H4 J

1"1""
0.1%

1-0. f=15kHz 1

~
+

I

U

1%

~

10

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

10

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
OUTPUT POWER

FREQUENCY

vs

10%

Iz

10%
f= 15kHz

+

c

~

1%

i

-

II'toI.lI

I
+

f=~~W"

J
IIf

I

I

~

r--I'-

""'"

0.1%

RL=SO
AV=-2VN
BTL

1%

Jj

f=2OHz

~

Ill!
0.1 "A

If

z

~

~

~

I I 1111111

0.01%
0.01

0.1
Po - OUtput Power - W

~
PO=1.0 W~

~

RL=40
AV =-24 VN
BTL

i!:

~ I/'

V" 1--.

0.01 %
20

10

100

I
PO=0.5W
1k

10k 20k

f - Frequency - Hz

Figure 14

Figure 13
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY

OUTPUT POWER

vs

10%

10%

PO=1W
RL=SO
BTL

=
~

I
I "

b
AV=-24VN

%

/

L

AV=-12VN

0.1,...

1\ j/

If

~

~

~

1-"""

0.01 %
20

V

I,;' AV=-2VN

ill

AV=~VN

~

1k
f - Frequency - Hz

1%

i

l""- I'--

f=15kHz

c0

!!

•
:z:

r-I'-

Iz

t-!,= 1 kHz

0.1%

I

~

+

S;;

f=2OHz

Q

V
100

+

u

v

./

~

t- BTL

c

~

F RL=SO

~ AV=-2VN

z=

'0

+

j

A

Po = O.25W

111111

i!:
10k 20k

0.01%
0.01

111111
0.1
Po - OUtput Power - W

10

Figure 16

Figure 15

~1EXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-357

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S204B

MAY 1999 REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

va

TOTAL HARMONIC DISTORTION PLUS NOISE

va

OUTPUT POWER
10%

~

I

I
I
'z7

10%
RL=sO
Av=-6VN

f!;:: I'"--~L

+

ii

~

OUTPUT POWER

......... ~15kHZ

I

......... f = 15 kHz
.... r-.
1%

1%

f='1 kHz
r-.!,=1 kHz
0.1%

i"- t---.

'"""I ItH4

'""

IT

I f~11~~

0.1%

f=2OHz

6
i!:

t- RL=SO

t- AV=-12VN
BTL

0.01%
0.01

0.01%
0.01

0.1

Po - Output Power - W

Figure 17

Figure 18

TOTAL HARMONIC DISTORTION PLUS NOISE

va

TOTAL HARMONIC DISTORTION PLUS NOISE

va

OUTPUT POWER

FREQUENCY
10%

10%

+

~

1%

r-

oS!

c

~

J
~

1%

f = 1 kHz

Q

ii

RL=320
Av=-1 VN
SE

....f = 15 kHz
rr
II

••zc
I

I.......

0.1%

Po =25mW ~

.' IJIII

f=2OHz

P""

i"f...

0.1%
Po =50mW

I

Z
+
Q

:z:

I-

10

0.1

Po - Output Power - W

~

- RL=SO
- AV=-24VN
BTL
0.01%
0.01

jOiliiml-

/
0.01%
20

0.1
Po - Output Power - W

T

100

1k
f - Frequency - Hz

Figure 19 '

Figure 20

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA0112
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%

I

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
100/0

F RL=32n
AV=-1 VN
t- SE

+

~

z

+

c

I

1%

i
j

RL=10kn
AV= 1 VN
SE

::
'0

t-

1%

2i

.2
c

.~

f= 15 kHz

o 0.1%

~

Ii

-

J:

!

0.1%

'7

I

z

!

fJ 1 JHl

z+

f=20Hz

~

Q

-.l..:o::t!

i!:

0.01%
0.01

Vo=1 VRMS

'70.01%
J:
~

III

0.001%
20

0.1
Po - Output Power - W

100

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT VOLTAGE

OUTPUT NOISE VOLTAGE
vs
BANDWIDTH

100/0

~
+
l5

;:

100

RL=10kn
AV= 1 VN
SE

VDO =

>::l.

1%

~

~

~

80

I
GI
CI

0.1%

5\1

90 I-R =4n

i

I

10k 20k

Figure 22

Figure 21

::

1k
f - Frequency - Hz

.~

f= 20 Hz

.........

z
'5
a.
'5

f= 15kH~

0

~~

If 0.01%

~

I

70
60
AV=-24VN

50

AJI~ _12 Vl r V
1

40

1111111

30

AV=-6VN

c

>

i!:

20
1-"''''

10

0.001%
0.1

3

o

....

Vo - Output Voltage - VRMS

V
r

AV=-2VN

~

10

~~

~ .... 1-:::

/

100

1k

10k

BW - Bandwidth - Hz

Figure 23

Figure 24

-!II

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-359

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SEmNGS
SLOS204S - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY

SUPPLY RIPPLE REJECTION RATIO
vs
FREQUENCY

o

'B

-20

I

I

I
;

j
II)

o

RL=8C'
CB =0.47 I1F
BTL

1 1111111

t""

I I lUll

1e

AV=-1 VN ~
~

v

-80

v. -100

100

1k

-120
20

10k 20k

100

1k

f - Frequency - Hz

f - Frequency - Hz

Figure 25

Figure 26

CROSSTALK
vs
FREQUENCY

CROSSTALK
vs
FREQUENCY

10k 20k

0
PO=1W
RL=80
AV=-2VN
BTL

-20

-40
GI
'1:1
I

-60

0

.."..

-80
LEFT TO RIGHT.-100
-120
20

"",

..... 1"-

J

0

GI
'1:1
I

........

-60

;

-100

-20

-40

AV =-2 VN

-80

-120
20

..... 1"-

I
a
ia:

-

AV=-24VN

-60

-20

GI
'1:1
I

I

-40

RL=32'n
CB=OA7J1F
SE

Jill

RI~.rr TO LlSFr100

i--"""

1e

-40

-80

I--LEFT TO RIGHT

0

V

-80

10k 20k

-120
20

f - Frequency - Hz

RI~~~6~~~
I I 1111
100

1k
f - Freqilency - Hz

Figure 27

Figure 28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 855303 • DAllAS, TEXAS 75265

~

...J

-100

1/

1k

PO=1W
RL=80
AV=-24VN
BTL

10k 20k

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

TYPICAL

CHARACTERiSTiCS

CROSSTALK

SHUTDOWN ATTENUATION

vs

vs

FREQUENCY

FREQUENCY

0

0
VO=1 VRMS
RL=10n
AV=-1 VN
SE

-20

III
'a
I

...
I.

VI = 1 VRMS

II

-20

I
RL = 10 kn, SE

-40

-40

III
'a
I

c

i

~o

~

::I
C

e

(J

-80

~~

-100

-120
20

....

/

1-"'1-'
~ i-'

-

-80

/"

.........

r-

-100

~=8n,mi(

RIGHT TO LEFT

I I 1111111

100

RL=32n,SE

!

LEFT TO RIGHT/-

1k
f - Frequency - Hz

-120
20

10k 20k

1k

100

10k 20k

f - Frequency - Hz

Figure 29

Figure 30
SIGNAL-TQ-NOISE RATIO

vs
FREQUENCY
140
PO=1 W
RL=8n
BTL

130
III
'a
I

j
CD

120 !!!;:

........
110

.!!
0

z
~

ic

100

~

90

a:

80

I

z

II

AV=~VN

AV=-2VN

I

r--

/

III

~
K .....
AV=-24VN

r-..;;

1\

r--::::

Av=-12VN -

II)

70

60
20

1k

100

10k 20k

f - Frequency - Hz

Figure 31

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-361

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS

SL0S204B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE

10

360°

.ll!!l'11

7.5

i"I

5

m

2;5

......

'V

I

c

'iii

270°

Phase

0

180°

-2.5
RL=8n
Ay=-2YN
BTL

-0

90°

11111

-7.5

-10 11111111 II
10
100

1k

10k

100k

1M

0°

f - Frequency - Hz

Figure 32

CLOSED LOOP RESPONSE
360°

30
25

270°

20
GaIn

m

15

'V

I

~

IJ~~~

,;r<

10

'\

5

o

RL=8n
Ay =-6 YN
BTL

90°

111111

-10
10

11I11111 II
100

1k

10k
f - Frequency - Hz

100k

Figure 33

~TEXAS

3-362

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

J
1:1.

~

CJ

1M

0°

TPA0112
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S204B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE

30

360°

IIIIII!

25

Gain
270°

20
ID

"I

i

15

)'1
I

\

Phase

""

10

.... 1"-

,

5

o
-5

RL=80
AV =-12 VN
BTL
J

111111

11111111 II
100

-10
10

J

180°

~

1k

10k

100k

1M

o·

f - Frequency - Hz

Figure 34

CLOSED LOOP RESPONSE

30

Galn

,

25

20
ID

"I

i

7~
15

.....

Phase

10

I'

....

,

5

o
-5

-10
10

RL=80
AV =-24 VN
BTL
111111

I

11111111 II
100

180° •

it

90°
II

1k

10k
f - Frequency - Hz

100k

Figure 35

~TEXAS

INSTRUMENTS
POST OFFICE BOX 656303 • DAUAS. lEXAS 75265

1M

0°

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETIINGS
SLOS204B- MAY 1999- REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

VB

VB

LOAD RESISTANCE

LOAD RESISTANCE

3.5
3
~

lJ

2.5
2

~

1.5

,

~I

1\

'5

1000

J

10%THD+N

'5

I

rP

0

J:>

"

1%THD+N~ r--..

0.5

o

I

8

750

~

~

0

o

AV=-1 VN
SE
1250

I

I

1500

AV =-2 VN
BTL

~ 10%THD+N

500

~~

250

16
24
32
40
48
RL - Load Resistance - 0

56

1%TH~~

o

64

1

o

8

16
24
32
40
48
RL - Load Resistance - 0

Figure 36

VB

OUTPUT POWER

OUTPUT POWER

I

c

L

i

1.2

i

0.8

iL V"
JL

I

0.6

IL

j.

a..
Q

a..

0.4

rL

--

~

0.35
~
I

0.3

i

0.25

I
I

0.15

Q

a..
1=1 kHz
BTL
Each Channel

1.5
Po - Output Power - W

2

/

0.1

1/

V

-

40-

K

............

l'-

r--.. ~ I
o
o

.......

"80

f=1 kHz
SE
Each Channel

320

0.05

2.5

r--.I-.,

""""

V
j

0.2

I

8~

0.5

/

c

40

0.2

o
o

0.4

I
13~-

~

L

1.4

64

POWER DISSIPATION

VB
1.8

~

56

Figure 37

POWER DISSIPATION

1.6

~

I

~

~

~

M

~

~

Po - Output Power - W

Figure 38

Figure 39

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

M

M

TPA0112
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7

\

8JA4

6

3:
I

c

5

'iii

4

i

OJ

I

1

3

0

11.

8JA1,2

I
Q

11.

2

o

11-

_!.

1\
1\

"- ~

jJA3

is

!

8JA1 = 45.9°CIW
8JA2 = 45.2°CIW _
8JA3 =31.2°CIW
8JA4 = 18.6°CIW

"""'" i'....

~
f'

\
~

~ ~1\

"'" ~

~o~o

"

0
~
@ ~ ~ 1001~1@1~
TA - Ambient Temperature - °C

Figure 40

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-365

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS

SLOS204B - MAY 1999 - REVISED MARCH 2000

THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 41)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

SIde VIew Ca)

Thermal
Pad

End View Cb)

Bottom View Ce)

Figure 41. Views of Thermally Enhanced PWP Package

APPLICATION INFORMATION
selection of components
Figure 42 and Figure 43 are a schematic diagrams of typical notebook computer application circuits.

~1ExAs

INSTRUMENTS
3-366

POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

Right CIRHP
Head- 0.47!J.F
phone
Input
20
Signal

--J

23

8
CRIN
0.47 !J.F

RHPIN
RLINEIN

R
MUX

ROUT+

21

ROUT-

16

RIN

T

-=- 14
PC BEEP
Input ---1f-!.:!...j!-,-,=""""-I
Signal CPCB
0.47 !J.F"..17'-+_ _ _-I

VDD

1 kQ

100kQ

2
3

GAINO
GAIN1

15

SElBTL

Galnl
MUX
Control

PVDD

CSR

Depop
Circuitry

-:J' 0.1 !J.F
Power

Left CILHP
Head- 0.47!J.F
phone
Input
Signal

-1

18 See Note A

1-.!....!..91-'''---.-- VDD
-

VDD

Management !-'B=-:Y;:P=A==S=S+-1:..:.1_--.
SHUTDOWN 22

VDD

CSR
0.1!J.F

I~"--'WrlHJ:v;;;::;::=~.J----,G=N.::Dll

19

"'I'

CBYP

-:J'

To
0.47!J.F
SystemControl

LOUT+

4

LOUT-

9

112
13,24

1 kQ

COUTR
330!J.F

LIN

100kQ

NOTE A.

A 0.1 !J.F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 !J.F or greater should be placed near the audio power amplifier.

Figure 42. Typical TPA0112 Application Circuit Using Single-Ended Inputs and Input MUX

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TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

ROUT+

21

COUTR
33OI1F
ROUT-

16

VDD

1 len

100 len

~~~~L-~:::711~~~~~~==~~I---=-=-=t---''"''--....
PVDD 18 SHNomA
VDD

r-

Depop
Circuitry
Power
Management

L

t--'I/\IIr-.....__

VDD

19

BYPASS
SHUT-

11

DOWN

22

~:;:::;:::=:;-rJ--'=ll

CSR
1::'0.1I1F

-

VDD

T

CSR
0.111F
CBYP

To 1::' 0.4711F
System-

1 kO

Control
LOUT+

COUTR
330I1F

LIN

LOUT-

9

100 len
NOTE A.

A 0.1 I1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals. a larger
electrolytic capacitor of 10 I1F or greater should be placed near the audio power amplifier.

Figure 43.1Yplcal TPA0112 Application Circuit Using Differential Inputs

~lExAs

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POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
gain setting via GAINO and GAIN1 Inputs
The gain of the TPA0112 is set by two input terminals, GAl NO and GAIN1.
Table 1. Gain Settings

GAINO
0
0
1
1
X

GAIN1
0
1
0
1
X

SElBTL

0
0
0
0
1

Ay
-2VN
-6VN
-12VN
-24VN
-1 VN

The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, ZI, to be dependant on the gain setting. The actual gain settings are controlled
by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input
impedance will shift by 30% due to shifts in the actual resistance of the input impedance.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 10 kn, which is the absolute minimum input impedance of the TPA0112. At the higher gain
settings, the input impedance could increase as high as 115 kn.
input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r-----------s~~=: ---11f---4I--..::.IN'--+1~~
I

"I
-=-

ZF

I
I

The input impedance at each gain setting is given in the table below:

Ay
-24VN
-12VN
-6VN
-2VN

ZI
141<0
261<0
45.51<0
911<0

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TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
The -3 dB frequency can be calculated using equation 1:
f
1
-3 dB - 21t C(R II RI)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.

input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a
high-pass filter with the corner frequency determined in equation 2.

fC(highpass) =

(2)

21t~,C,

The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where Z, is 710 k.Q and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.

C 1
, - 21tZ, fc

(3)

In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 IlF. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capaCitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.

~TEXAS

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TPA0112
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
power supply decoupling, Cs
The TPA0112 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 IlF placed as close as possible to the device VDD lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capaCitor of 10 IlF or greater placed near
the audio power amplifier is recommended.

midrail bypass capacitor, CBYP
The midrail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the mid rail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CBYP, values of 0.47 IlF to 1 IlF ceramic or tantalum low-ESR capacitors are recommended
for the best THO and noise performance.

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capaCitor (Cd is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

(4)

fC(hlgh)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 IlF is chosen and loads vary from 3 n,
4 n, 8 n, 32 n, 10 kQ, to 47 kQ. Table 2 summarizes the frequency response characteristics of each
configuration.

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TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode

RL

Cc

Lowest Frequency

3n

330 I1F

161 Hz

4n

33Ol1F

120Hz
60Hz

an

33OI1F

32n

33OI1F

15Hz

10,ooon

330 I1F

0.05.Hz

47,ooon

330l1F

0.01 Hz

As Table 2 indicates, most of the bass response is attenuated into a 4-0 load, an 8-0 load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESA capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capaCitor in the circuit. The lower the equivalent value of this
resistance the more the real capaCitor behaves like an ideal capacitor.

bridged-tied load versus slngle-ended mode
Figure 44 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0112 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 5).

v

_ VO(PPJ

(rms) -

(5)

212

2
V(rms)

Power = - RL

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TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

voo

J' ;

Voo

RL

J'!

vO(PP)

2x vO(PP)

Figure 44. Bridge-Tied Load Configuration

In a typical computer sound channel operating at 5 V, bridging raises the power into an s-n speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 45. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33j.1F to 1000 j.1F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
fc =

(6)

1

2nRL C c

For example, a 6S-j.1F capacitor with an s-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

voo

~dB~----~~====

Figure 45. Single-Ended Configuration and Frequency Response

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/

TPA0112
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

single-ended operation
In SE mode (see Figure 44 and Figure 45), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier'S gain to 1 VN.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The internal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 46).
100

,/

--fV'VVVffll-

V(LRMS)

IOO(avg)

Figure 46. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In anSE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

"'TEXAS

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TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Efficiency of a BTL amplifier

P

=

~
SUP

(7)

Where:
VLrms2
Vp
PL = ~' andV LRMS = 12' therefore, PL

=

Vp 2
2RL

J"

1
Vp .
_ 1
V
11:
2Vp
looavg = it 0 RL sJn(t) dt - it x R: [cos(t)] 0 = 11: RL

and
Therefore,
_ 2 Voo Vp
Psup 11: RL

substituting PL and Psup into equation 7,
V p2

Efficiency of a BTL amplifier
Where:

PL =Power devilered to load
Psup =Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL =Load resistance
Vp = Peak voltage on BTL load
looavg =Average current drawn from
the power supply
Voo =Power supply voltage
llBTL =Efficiency of a BTL amplifier

~

11:Vp
2Voo Vp = 4 Voo
11: RL

Therefore,

_11:~
T]BTL -

(8)

4 Voo

Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-n loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.

Table 3. Efficiency Vs Output Power in 5-V 8-n BTL Systems
Output Power

Efficiency
(%)

Peak Voltage
(V)

Internal Dissipation

(W)

0.25

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.47t

0.53

(W)

t High peak voltages cause the THO to Increase.
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.

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TPA0112
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature the internal
dissipated power at the average output power level must be used. From the TPA0112 data sheet, one can see
that when the TPA0112 is operating from a 5-V supply into a 3-0. speaker that 4 W peaks are available.
Converting watts to dB:

P

PdB = 10Log-.lOC = 10Log 41 Ww = 6 dB

(9)

P ref

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
15 dB = -9 dB (15 dB crest factor)
12 dB = -6 dB (12 dB crest factor)
9 dB = -3 dB (9 dB crest factor)
6 dB = 0 dB (6 dB crest factor)
3 dB 3 dB (3 dB crest factor)

6 dB 6 dB 6 dB 6 dB 6 dB -

=

Converting dB ba9k into watts:
Pw

=

10PdB/10 x P ref

=
=
=

63 mW (18 dB crest factor)

(10)

125mW (15 dB crest factor)
250 mW (9 dB crest factor)

= 500 mW (6 dB crest factor)
= 1000 mW (3 dB crest factor)
= 2000 mW (15 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-0. system, the internal dissipation in the TPA0112 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0112 Power Rating, 5-V, 3-0., Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

4

2W(3dB)

1.7

-3°C

4

1000 mW (6 dB)

1.6

6°C

4

500 mW (9 dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

78°C

4

63 mW (18 dB)

0.6

96°C

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2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

crest factor and thermal considerations (continued)
Table 5. TPA0112 Power Rating, 5-V, &-0, Stereo
PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/ehannel)

MAXIMUM AMBIENT
TEMPERATURE

2.5W

1250 mW (3 dB crest factor)

0.55

1000 e

2.5W

1000 mW (4 dB crest factor)

0.62

94°e

2.5W

500 mW (7 dB crest factor)

0.59

97°e

2.5W

250 mW (10 dB crest factor)

0.53

102°e

The maximum dissipated power, POmax, is reached at a much lower output power level for an 8 0 load than for
a 3 0 load. As a result, this simple formula for calculating POmax may be used for an 8 0 application:

2Vbo
POmax = n;2R

(11 )

L

However, in the case of a 3 0 load, the POmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 0 load.
.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to 9JA:
9 JA

=

1
Derating Factor

=

_1_
0.022

= 45°C/W

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given 9JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0112 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T A Max = T J Max - 9 JA Po

(13)

= 150 - 45(0.6 x 2) = 96°C (15 dB crest factor)
NOTE:
Intemal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.

Tables 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0112 is designed with thermal protection that tums the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-0 speakers dramatically increases the thermal performance by increasing amplifier
efficiency.

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2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS204B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

SE/BTL operation
The ability of the TPA0112 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0112, two separate amplifiers drive OUT+ and OUT-. The SE/BTL input (terminal 15) controls
the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When SE/BTL is held
low, the amplifier is on and the TPA0112 is in the BTL mode. When SE/BTL is held high, the OUT-amplifiers
are in a high output impedance state, which configures the TPA0112 as an SE driver from LOUT+ and ROUT+
(terminals 4 and 21).100 is reduced by approximately one-half in SE mode. Control of the SElBTL input can
be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 47.

20

RHPIN

23

RLiNEIN

R

MUX
ROUT+

8

21

RIN

Voo
ROUT-

16
100kn

SEiSTL

15 100 kn

~

n

,----~
Figure 47. TPA0112 Resistor Divider Network Circuit

Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-kO/1-1--+-------

ROUT+

>-.....---1-----

ROUT-

-----------t---e

----t PC1 ...._B_ee_p~
Power
Management

GAINO
GAIN1
SElBTL

PVDD
VDD
BYPASS
SHUTDOWN

' - - - - - GND

LHPIN---I
LLINEIN - - - - I

UN

>--+---1-----

LOUT+

>-.....-------

LOUT-

- - - - - - - - - - + -____

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TPA0122

2·W STEREO AUDIO POWER AMPLIFIER

WITH FOUR SELECTABLE GAIN SETTINGS
SL0S247B - JUNE 1999 - REVISED MARCH 2000

AVAiLAtii-E ui'TiuiliS

PACKAGED DEVICE
TA

TSSOP't
(PWP)

-40°C to 85°C

TPAOI22PWP

t The PWP package is available taped and
reeled. To order a taped and reeled part. add
the suffix R to the part number (e.g .•
TPAOI22PWPR).

Terminal Functions
TERMINAL
NAME
NO.

I/O

DESCRIPTION

BYPASS

11

GAINO

2

I

Tap to voltage divider for internal mid-supply bias generator
Bit 0 of gain control

GAINI

3

I

Bit 1 of gain control

GNO

1.12.
13.24

Ground connection for circuitry. Connected to the thermal pad

LHPIN

6

I

Left channel headphone input. selected when SElBTL is held high

LIN

10

I

Common left input for fully differential Input. AC ground for single-ended inputs

LLiNEIN

5

I

Left channel line Input. selected when SElBTL Is held low

LOUT+

4

0

Left channel positive output in BTL mode and positive output in SE mode

LOUT-

9

0

Left channel negative output in BTL mode and high-Impedance in SE mode

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > I-V (peak-ta-peak) square wave is input
to PC-BEEP or PCB ENABLE Is high.

PCB ENABLE

17

I

If this terminal is high. the detection circuitry for PC-BEEP is overridden and passes PC-BEEP through
the amplifier. regardless of its amplitude. If PCB ENABLE is floating or low. the amplifier continues to
operate normally.

7.18

I

Power supply for output stage

20

I

Right channel headphone Input. selected when SElBTL is held high

8

I

Common right input for fully differential Input. AC ground for single-ended inputs

PVOO
RHPIN
RIN
RLiNEIN

23

I

Right channel line input. selected when SElBTL Is held low

ROUT+

21

0

Right channel positive output in BTL mode and positive output in SE mode

ROUT-

16

0

Right channel negative output in BTL mode and high-Impedance in SE mode

SHUTOOWN

22

I

Places entire IC In shutdown mode when held low. except PC-BEEP remains active

SElBTL

15

I

Input MUX control input. When this terminal is held high. the LHPIN or RHPIN and SE output is selected.
When this terminal is held low. the LLiNEIN or RLiNEIN and BTL output are selected.

VOO

19

I

Analog VOO input supply. This terminal needs to be isolated from PVOO to achieve highest performance.

="TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-383

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, VDD ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to VDD +0.3 V
Continuous total power dissipation ..................... intemally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

DERATING FACTOR
2.7W=I=

PWP
=1=

1.7W

1.4W

Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
High-level input voltage, VIH

MIN

MAX

4.5

5.5

SElBTL

4

SHUTOOWN

2

SHUTDOWN

0.8
-40

Operating free-air temperature, TA

V
V

3

SElBTL

LOW-level input voltage, VIL

UNIT

85

V
°C

electrical characteristics at specified free-air temperature, Voo = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

IVool

Output offset voltage (measured differentially)

VI=O,

PSRR

Power supply rejection ratio

VOD =4.9Vt05.1 V

IIIHI

High-level input current

IIILI

Low-level input current

IDO

Supply current

IOD(SD)

Supply current, shutdown mode

TYP

MAX

UNIT

25

mV

VDD=5.5V,
VI=VDD

900

nA

VOO=5.5V,
VI=OV

900

nA

77

BTL mode

18

SEmode

9
150

~TEXAS

3-384

MIN

AV=2

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

dB

rnA
300

I1A

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

operating characierisiiC8, VDD

=5 V, iA =25~C, fiL =S n, Gaiii .. -2 V/V, BTL rnQd~

PARAMETER

TEST CONDmONS
THO=1%,
RL=4Cl

1= 1 kHz,
1 = 20 Hz to 15 kHz

Po

Output power

THO+N

Total harmonic distortion plus noise

PO=1 W,

BOM

Maximum output power bandwidth

THO=5%

Supply ripple rejection ratio

1=1 kHz,
CB=0.47 ILF

SNR
Vn

Z,

I

BTL mode

Signal-to-noise ratio
Noise output voltage

CB = 0.47 ILF,
1 = 20 Hz to 20 kHz

I BTL mode
I SE mode

MIN

TYP

MAX

UNIT

1.9

W

0.5%
>15

kHz

68

dB

105

dB

16
30

ILVRMS

See Table 1

Input impedance

TYPICAL CHARACTERISTICS

Table of Graphs
FIGURE
vs Output power

1,4-7,10-14,
16-19,21

vs Frequency

2,3,8,9, 14,
15,20,22

THO+N

Total harmonic distortion plus nOise

vs Output voltage

23

Vn

Output nOise voltage

vs Bandwidth

24

Supply ripple rejection retio

vs Frequency

25,26

Crosstalk

vs Frequency

27-29

Shutdown attenuation

vs Frequency

30

Signal-to-nolse ratio

vs Frequency

SNR

Closed loop response
Po
Po

Output power
Power dissipation

31
32-35

vs Load resistance

36,37

vs Output power

38,39

vs Ambient temperature

40

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • OAllAS, TEXAS 75265

3-385

TPA0122
2·WSTEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAJN SETTINGS
SLOS247B

JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

10%

10%
AV=2VN
f=l kHz
BTL

~
Z
+

J

c

~

/

I I

J
RL=40

1%

~Q

=
~

oS!
c
0

..

!

J:

OJ 0.1%

I

..L

Iill

I

RL=80

.L

RL=30

II

--.

I

1==
=
I--

Av=-12VN

./

....

~

~ V-

f'.

""

~I

'\

""

AV =-24 VN

L IL

l

...

PO=1.75W
RL=30
BTL

AV=-2VN

I'

IH

Z

+

AV =-6 VN

Q

j!:

1111

I I II

Iill

II

~

0.01%
0.5 0.75

1

1.25 1.5 1.75 2

2.25 2.5 2.75

0.01%
20

3

100

Po - Output Power - W

Figure 1

lk
f - Frequency - Hz

10k 20k

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

RL=30
AV= 2VN
BTL

·1z

+

k"

c

0

i!
0

~

1%

r- I-i'""

f = 15 kHz

J

..

Po =1.0W

V

Po =0.5W

"

·s0
!

.

./~

~

V

-

J:

i

~I

to-

0.1%

0.01 %
20

II
100

III
1111

+

)

...J:

I I I Lil
lk
f - Frequency - Hz

10k 20k

0.01%
0.01

Figure 3

t-"

1111

~TEXAS

POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

RL=30
AV=-2VN
BTL

0.1
Po - Output Power - W

Figure 4

INSTRUMENTS
3-386

J

f=20Hz

Z
Q

Po =1.75W

f=lkHz

--w

10

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS·
SLOS247B - JUNE 1999 - REVISED MARCH 2000

_ ... _.- •• _ ••• -"'. "".. __ .""... .fI"IIo"
I 'I""n..AL \"nAnA\" I ':;"1;:) I I\";:)
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10%
CD

f = 15 kHz

1/1

·0
Z

+

,= 15 kHz

0

Iii{

c

'E0

~u

1%

-

·c0

i

::c

Ii
~I

0.1%

I

~I

I

fl= 11

1%

'=1 kHz

t-

r-.... t--..

,=1201Hl

T

k~l T

..,.......
f~2~~~1

r-......

-4J..1

II

r-t-

0.1%

Z

+

Q

t- RL=30

::c

I-

t0.01%
0.01

t-

Av=~VN

RL=30

t- Av=-12VN

BTL

0.01%
0.01

0.1
Po - Output Power - W

BTL

Figure 6

Figure 5
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY
10%

10%
f = 15 kHz

CD

1/1

"0

z

z

1/

+
c

0

~u

1%

i

.2

""'......."

fo"""-

0.1%

0.1

Ii
~

""

I

Z

+

+

Q

t- RL=30

0.01%
0.01

::c

AV=-24VN
BTL

I-

0.1
Po - Output Power - W

10

~ ....

t-

::c

t-

1/

./

E
01

Z

I-

AV=-12VN

0

f=20Hz

::c

Q

r-

c

E
01

::c

I;'

Av =-24 VN

1%

Q

0

~I

.

~

f = 1 kHz

.......

"2

Ii

PO=1.5W
RL=40
BTL

.~0

I

+
c

'E0

10

0.1
Po - Output Power - W

r0.01%
20

-

./

V
AV=-2VN

-~

Av=~VN

IIIIIII I

100

1k

10k 20k

f - Frequency - Hz

Figure 7

FigureS

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-387

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S247B - JUNE 1999 - REVISED MARCH 2000

.

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
va
FREQUENCY
10%
RL=4n
Ay=-2YN
BTL

Iz

1

+

c

~

I
+

I

c

1%

~

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%
RL=4n
Ay =-2 YN
BTL

~

1%

!
is

PO=1.5W

Q

.!:!
c

r- ....

.!:!

0

..~

I~~

I!

0.1

~

'*~

~~

Po = 0.25 W

0
i!:

,..

0.01%

100

20

1k

10k 20k

f=1 kHz
I I 11111

to--

!
I--

~

1-0

:z:

PO=1.0 wi==

I

Z

0.1%

~
I

Z

f=2OHz

0
i!:
0.01%
0.01

0.1

Figure 10
TOTAL HARMONIC DISTORnON PLUS NOISE
vs
OUTPUT POWER

10% _ _

10%~~1I

-

+

r-

f=15kHz

I"-r--t--

1%

I
+

f=2OHz
RL=4n
t- Ay =45 YN
r BTL

i;

~

0.01%
0.01

J
If
~

~--I-~~~~-+++Hffi

I I 11111
0.1
Po - Output Power -

w

_ff--+-t-t-t1Ht1

i -..

~~

0.1% ~

1-~-+--I-R"I''I+Wo=-.
f= 15 kHz ..
-r-.
1

~iS: 1%~I§~II~~~~J~~!I!I
E

I

f=1 kHz
I 1111

r-

~

~

10

f=1kHz

r-,!.1!t,l;1 r

0.1%~¥~1~!~11~!11
E

RL=4n
Ay -12 YN -H-H+tttt---+--H-t+1Ht1
BTL
0.01% L--J....J..
L.l..l..I,u
II.J.L
1111Iu--1...I-..L-'-'-J..J.l.LI.---"-'-...........~.
0.01
0.1
10

=

Po - Output Power -

Figure 11

Figure 12

~TEXAS

3--'388

10

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

I

!If

llrll

Po - Output Power - W

Figure 9

J

/

r--L I I IIIJ

f - Frequency - Hz

I

I

f = 15 kHz

INSTRUMENTS
POST OFFICE BOX 65S303 • DALLAS, TEXAS 75265

w

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S247B - JUNE 1999 - REVISED MARCH 2000

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
OUTPUT POWER

vs
FREQUENCY

10%

10%

iz

f= 15 kHz

,

.....

+

~

I

c

1%

!

~

I

I-"

0

I

f= 20 Hz

....... r--I--

]

0.1%

~I

1%

I

f= 1 kHz

Q

.!:!
c

II

+

15

I""

~

RL=SO
AV=4VN
BTL

J!

Po = 0.25W

~

0.1%

PO=1.0W -

J
If 0.01%

Z

+

j!:

PO=0.5W

j!:

I I 1111111

0.01%
0.01

.... ~

~

RL=40
AV =-24 VN
BTL

Q

0.1
Po - OUtput Power - W

III

111111

0.001%
20

10

100

1k
f - Frequency - Hz

Figure 13

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY.

vs
OUTPUT POWER

10%

I
.!:!

15

10%

PO=1W
RL=SO
BTL

+

I

1%

1J 11111

=t=

~

AV=-24VN

/

V

~

I
]

~

7z 0.01

I...

~

+

c

i

-"
1/

'"

1%

f= 15 kHz

fl=ik~1

.~
o 0.1%

AV=4VN :

I

l'

IIf 0.01%

AV=-8VN

f=20Hz
~

~

~

j!:

0.001 %
20

RL=SO
AV=4VN
BTL

J!

t- AV=-12VN
0.1%

10k 20k

Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

I

,l

100

10k 20.

1k

0.001%
0.01

f - Frequency - Hz

Figure 15

0.1
Po - Output Power - W

10

Figure 16

~TEXAS

INSTRUMENTS
POST OFF1CE BOX 655303 • DALlAS. TEXAS 75265

3-389

TPA0122
2·WSTEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS2478 - JUNE 1999

REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER
10%

f=:=

RL=8Q
~ AV=-{SVN
f-- BTL

:ll
'0
z

+
c

~

1%

.1c

r-- I-~

-

I II
f = 15 kHz

---

l-

f = 15 kHz

1%

,g
c

1=1 kHz

~

0

Ii

r- r-

:I:

J.I I III

f=1 kHz

~llill

]j 0.1%
~
I

If}2~

'"""I'-I"'"
0.1%

f=20Hz

Z

+

~

C
:I:

r- RL=8Q

I"'"

t-

0.01%
0.01

r-

0.1
Po - Output Power -

AV=-12VN
BTL

0.01%
0.01

w

0.1
Po - Output Power - W

Figure 17

Figure 18

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
10%

10%

.!z

-

+
c

~

1%

!c

......

I II

Ii

,....

i

Ii

:I:

Po =25mW -

,g

5

f~Jolrzr

0.1%

j

t'---

]j 0.1%
~
I

1%

is

~

I

SE

+

f= 1 kHz

1"""0.

RL=32Q
AV= 1 VN

I

f= 15 kHz

u

'c0

..4-

F

]j
~

10.01%

z

Z
+
C
:I:
t-

RL=8Q
t- AV=-24VN
BTL

0.01%
0.01

Po =75mW

ill

~

100

l..oIII
Po =50mW ~

~

r

i!:

ilL

0.1
Po - Output Power - W

10

0.001%20

Figure 19

IIIIII
100

1k
f - Frequency - Hz

Figure 20

~TEXAS

INSTRUMENTS
3-390

10

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS

----- .. -... -. ""..... _.......""....

SL0S247B - JUNE 1999 - REVISED MARCH 2000

I Yt"1"'AL ",nAn,,,... I enl-=» II"-=»

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

10%

10%
RL=32Cl
Av=-1 VN
SE

j
i

+

RL= 10kn
AV=-1 VN
SE

j
+

I

1%

i
g
I

I--- f = 15 kHz

u

~o

0.1%

~

~

i!:

~f=20Hz

0.1%

I
~

I;;:;-- 1= 1 kHz

If 0.01%

1%

If

Vo=1 vRMS

0.01%

"""-

~

,

i!:

0.001%
0.01

0.001%
20

0.1

100

Figure 22

Figure 21
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT VOLTAGE

OUTPUT NOISE VOLTAGE
vs
BANDWIDTH

10%

100

RL=10kn
AV=-1 VN
SE

j
+
c

II

IL
II

>::I.

80

I
CD

CI

70

~

60

J

50

:\l!

\

£

f=~"HZ
~

~

If 0.01%
z

-......

'S
So
:s

...,~

f=15kHz

~~

~

0

I

f=1 kHz

o

0.2 0.4 0.6 0.8

1

1.2

Av=-24VN
I-II
Avl1ltlLJ

40
30 f--

C

>

~
0.001%

VDDI=5Y I
R=4Cl

90

...L

1%

0.1%

10k 20k

1k
f - Frequency - Hz

Po - Output Power - W

20
10

1.4 1.6 1.8

2

o

-

AV =1-6

~I-'

---

10

Vo - Output Voltage - vRMS

V
....:w

K

~
j..o

-'-

V
V

V

~ 1--"'1-'

~

......I1dL"";
Av =-2 VN
llL

100
1k
BW - Bandwidth - Hz

10k

Figure 24

Figure 23

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-391

TPA0122
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO

SUPPLY RIPPLE REJECTION RATIO

va

vs

FREQUENCY

FREQUENCY

o

0

m

RL=sn
CB=0.47 IlF
BTL

-20

'Q

m

-20

'Q

I

...... 1"-

I

i

~

-40

II:

c

CD

~o

l

......
ii:
......

.!!

-80

r--Ay=-24YN
i----' ~

"r-.,

~

"- i"'"

::I

-100

V

,.

i---- 1--"'"

r-

(I)

.........

i'ii"

AV=-1 VN

~o

'-

!

~

-ao

-100 H-H+tttt---t--HH-ttHt-H-tlHiiit--t

1k

10k 20k

-120.':--1-L..1..J.JJ.~--J.......I-JL...J..J..I..I..I;':_.L....J-L...Ju..J.I~---:=

20

100

1k

f - Frequency - Hz

f - Frequency - Hz

Figure 25

Figure 26

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

0
PO=1W
RL=Sn
AV=-2VN
BTL

-20

-20

PO=1 W
RL=sn
AV = -24 VN
BTL

-40

m
'Q
I
.00:

~

-80

OJ

e

V

0

-a0

-100

LEFTTORIG~

I-

-120 ...... u-.
20

RI~H~ +~ LI~~

vI-./

+-J.A"

J.~

100

1k
f - Frequency - Hz

10k 20k
f - Frequency - Hz

Figure 27

Figure 28

~TEXAS

INSTRUMENTS
3-392

V

~

II:

111111111
100

r---I'-o.

J

Ay =-2 VN

-120
20

-40

g

0

1$

RL=32n
CB=0.47 IlF
SE

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

TYPiCAL CHARACTERiSTiCS
CROSSTALK

SHUTDOWN ATTENUATION

vs

vs

FREQUENCY

FREQUENCY

0

-20

ID

0
Vo= 1 VRMS
RL=10Q
AV=-l VIV
SE

VI=l VRMS

RL = 10 kQ, SE

-40

ID

I

....

..

-40

'D
I

'D

j

IJlm

11

-20

c

0

-60

i:::I

-60

i RL=32Q,SE

C

f

(.)

LEFT TO RIGHT

-80

~

-

-80

...........
-100

-100
RIGHT TO LEFT

-120
20

I I I 111111
100

lk

-120
20

10k 20k

100

t18~.
lk

f - Frequency - Hz

f - Frequency - Hz

Figure 29

Figure 30

10k 20k

SIGNAL-TO-NOISE RATIO

vs
BANDWIDTH
120
115
ID

'D
I

i

110

"-

~~
'-,

IIAV=I_2~UIII I-- rIIII
r-....
...

Av=-12VIV

~ r....

...

105

'1=

100 - - AV=-2VIV

II:

'0
~

ic

CI

95

I
II:

90

iii

z

PO='';'w
RL=811
BTL

I--- ~

I'-~ ...

r....
t:-- r-r-.

f\ r....

Av=-6VN

U)

85
80
20

100
lk
BW - Bandwidth - Hz

""'-

:::-...

"
10k 20k

Figure 31

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-393

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS

SL0S247B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE
10

360°

III~JI

7.5
5 IL

ID

2.5

"a

I

j

270°

0

I'

Phase

180° •

f

1'0

-2.5
RL=80
Ay=-2YN
BTL

-5

90°

iJJ1jj11L LWJ

-7.5
-10
10

lJmlJl
100

~
1k
10k
100k
f - Frequency - Hz

1M

00
2M

Figure 32

CLOSED LOOP RESPONSE
30

360"

RL~'8'O
Ay=-6YN
BTL

25
20 t-

2700
Gain

D

1111111 II
Phase

t-

"

180° •

f

5 t-

o t-5

90°

:1

-10
10

100

1\

1k
10k
100k
f - Frequency - Hz

~

1M

Figure 33

~TEXAS

3-394

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2M

0°

TPA0122
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS

_..._-- -. _... _. ...............
~

SL0S247B - JUNE 1999 - REVISED MARCH 2000
,..~

• 'I"""AL "nAnA'" I E:nl;;) I I"'~

CLOSED LOOP RESPONSE

30

"~UI

25

r""\

20

270°

'j.
Phase

r--

i\

5
RL=80
AV=-12VN
BTL

o

111111111

1800

\

90°

\

1111111

11111111111111
-10
10
100
1k
10k
100k
f - Frequency - Hz

J

~III
1M

2M

0"

Figure 34

CLOSED LOOP RESPONSE
30

""I
Gal~""-

II
25

360°
RL=80
AV=-24VN

~TL
270°

20

rj
......

Phese
180°

I""

5

i\

o

-10
10

J

90°

~
100

1k
10k
100k
f - Frequency - Hz

"

1M

2M

0°

Figure 35

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-395

TPA0122
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999

REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER·
vs
LOAD RESISTANCE

OUTPUT POWER
vs
LOAD RESISTANCE

3.5

J

3
~

0
D.

2.5
2

\

0
D.

10% THD+N

:;

~
1%THD~~

0

I

rP
0.5

o

8

I

1000

;

\

1.5

o

~I

l\

:;

~

Ay=-1 YN
SE
1250

I

;

1500

Ay =-2 YN
BTL

750

~

0

b
D.

~ 10%THD+N

500

\~

250

"""

16
24
32
40
48
RL - Load Resistance - 0

56

o

64

1%TH';:~r-..
o

1

1

8

16
24
32
40
48
RL - Load Resistance - 0

Figure 36

POWER DISSIPATION
vs
OUTPUT POWER

1.8

0.4

1.6

0

ir::I.

1.4

LV

1.2

~

;

0
D.
I

0.8
0.6

Q

D.

0.4

~

L

I

c

iL

0.35
~
I

0.3

c
0

40

JL
IL

ir::I.

0.25

~

;

0.2

~

0.15

I

rL

o
o

r3~ _

--

~

Q

8-;-

D.

f= 1 kHz
BTL
Each Channel

0.2
0.5

1.5
Po - Output Power -

2

0.1
0.05

2.5

w

I

V

L ......

r-...... .... ......

1

'L
V

o
o

~

'"

320

f=1 kHz
SE
Each Channel

I
u

~

M

~

Po - Output Power -

Figure 39

~TEXAS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

K

"80

l'

INSTRUMENTS

40-

..........

~

Figure 38

3-396

64

Figure 37

POWER DISSIPATION
vs
OUTPUT POWER

~

56

M

w

U

~

TPA0122
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS2478 - JUNE 1999 - REVISED MARCH 2000

APPLiCATiON

ii~FORiviAiiOr~

POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7

\

8JA4

6
~
I

c

5

·iii

4

i

III

;

"-

8JA3

i5

3

JI

0

11.
I
Q

11.

......."

8JA1,2
2

o

1\
1\
l'..

~~

i'

8JA1 = 45.9°CIW
8JA2 = 45.2°CIW
8JA3 = 31.2°CIW
8JA4 = 18.6°CIW

\.
l\

"1\
~~
......."

~

~040

0

~

~

~

00

"

1001~1~1~

TA - Ambient Temperature - °C

Figure 40

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-397

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S247B-JUNE 1999- REVISED MARCH 2000

APPLICATION INFORMATION
0.47/lf

{r-1

-

LOUT+

Gain
Setting

I

2

0-=---=-

RLiNEIN

o----!- GAIN1

SHUTDOWN

OA7 11F
0.47 11F

II

LHP 0

6 LHPIN

Ii'

-=

I

LOUT-

1:0

LOUT+

Ij-----!- LLINEIN

0.4711F

1:0

GND

GAINO

4

LLINEO

GND

r: =i=

.-!8
9

0.47 11F

0.47 I1F

J

10
11
12

PVDD
RIN

ROUT+
RHPlN
VDD
PVDD
PCB ENABLE

LOUT-

ROUT-

LIN

SElBTL

BYPASS
GND

PC-BEEP
GND

I

~

0 RLiNE

22

SHUTDOWN

21

ROUT-

OA7/lf
20

II

0 RHP

19

I
18

0.111F

0.111F

17
16

T 1
T1011F~

I

13

il::-

GND

PCB ENABLE
ROUT-

15
14

VDD

-'" SElBTL

-I~

Pc-BEEP

0.47 11F

Figure 41.1Ypical TPA0122 Application Circuit

selection of components
Figure 42 and Figure 43 are a schematic diagrams of typical notebook computer application circuits.

-!!1TEXAS
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INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

Right CIRHP
Head- 0.47 ILF
phone
Input
Signal

-1

23

RliNEIN
ROUT+

8

21

RIN

CRIN
0.47ILF

T
-----j

-=PC BEEP
14
Input
Signal CPCB
0.47 !1F17

COUTR
330ILF

PC-BEEP

PC
ENABLE

ROUT-

PCBeep

16

VDD

11<0

1001<0
2
3

GAINO
GAIN1

Gain!
MUX
r-+-==-=-='--I Control

PVDD 18 See Note A
I---'-'-'=-JI--'-"--.-- VDD
Depop
Circuitry

VDD
Power
Management BYPASS

Left CILHP
Head- 0.47 ILF
phone
Input
Signal

--7

L

~M=::::-T--G=:N""Dll

.---'VI/\r-..........

19

CSR
-:J:'0.1 ILF
VDD
-

11

CBYP
To -:J:' 0.47 ILF
SystemControl

LOUT+

4

LOUT-

9

112,
13,24

11<0

COUTR
330ILF

LIN

CliN
0.47ILF

T

1001<0
NOTE A.

A 0.1 ILF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 ILF or greater should be placed near the audio power amplifier.

Figure 42. "TYpical TPA0122 Application Circuit Using Single-Ended Inputs and Input MUX

~TEXAS

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TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
N/C
20

RHPIN

CCRINRight 0.47 /LF
23
Negatlv";;r1
Differential
Input
Signal
CRIN+
Right 0.47 /LF
Positive
8
Differential
Input
al
S:PE
PC
EP
14
Input
Signal Cp B
0.47/LF 17

-j

ROUT+

21

ROUT-

16

PVDD

18

VDD
Power
Management BYPASS

19

RIN

---::-1

GAINO
GAlN1
SElBTL
Left CIIHP
Head- 0.47 /LF
phone
Input
Signal

See Note A

I---'-'-~I-'-"'------

Gainl
MUX
Control

Depop
Circuitry

--1

L

~M=~r-.-:G"'N;.:Dll

11

VDD
CSR
1='0.1/LF
VDD

'T

CSR
0.1 JLF

CBYP

LOUT+

To 1=' 0.47/LF
SystemControl
1,12,
4
13,24

LOUT-

9

_-'V'Vv-.........

LUNEIN

1 kU

UN
CUN
0.47/LF

T-=

100kU
NOTE A.

A 0.1 /LF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 /LF or greater should be placed near the audio power amplifier.

Figure 43. Typical TPA0122 Application Circuit Using Differential Inputs

3-400

-!!1
TEXAS
INSTRUMENTS
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TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS2478 - JUNE 1999 - REVISED MARCH 2000

gain setting via GAINO and GAIN1 inputs
The gain of the TPA0122 is set by two input terminals, GAl NO and GAIN1.
Table 1. Gain Settings
GAINO

GAIN1

SE/BTL

Av

0

0

0

0

1

0

-2VN
-{)VN

1

0

0

-12VN

1

1

0

-24VN

X

X

1

-1VN

The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, Z" to be dependant on the gain setting. The actual gain settings are controlled
by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input
impedance will shift by 30% due to shifts in the actual resistance of the input impedance.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 10 kn, which is the absolute minimum input impedance of the TPA0122. At the higher gain
settings, the input impedance could increase as high as 115 kn.
input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

,r-----------I

ZF

Input ----------'l f-------.>-----"-=--t-'-'vV'v-___--I
Signal ----------;
R

The input resistance at each gain setting is given in the table below:

AV

ZI

-24VN

14 kO

-12VN
-{)VN

45.5 kO

-2VN

91 kn

26kn

~ThXAS

INSTRUMENTS
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3-401

TPA0122
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS

SLOS247B-JUNE 1999- REVISED MARCH 2000

APPLICATION INFORMATION
The -3 dB frequency can be calculated using equation 1:
f

1
-3 dB - 21t C(R II RI)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.

Input capacitor, C.
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZI, form a
high-pass filter with the corner frequency determined in equation 2.

fe(highpass) =

(2)

21t~,C,

The value of CI is important to consider as it directly affects the bass (low frequency) perfonnance of the circuit.
Consider the example where ZI is 710 k.Q and the specification calls fora flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
C =_1_
,
21tZ, Ie

(3)

In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 J.1F. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tant~lum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.

~TEXAS

INSTRUMENTS
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TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

Ann, '''AT',.. ..' .MCnCllllATlnM

"'rr .... v"'.

IV"" 11"

",,1 .... _

...........

power supply decoupling, Cs
The TPA0122 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 J.lF placed as close as possible to the device Voo lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 J.lF or greater placed near
the audio power amplifier is recommended.

midrail bypass capacitor, CBYP
The mid rail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THO+N.
Bypass capacitor, CBYP, values of 0.47 J.lF to 1 J.lF ceramic or tantalum low-ESR capacitors are recommended
for the best THO and noise performance.

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cd is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

(4)

fC(high)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 J.lF is chosen and loads vary from 3 n,
4 n, 8 n, 32 n, 10 kn, to 47 kil. Table 2 summarizes the frequency response characteristics of each
configuration.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-403

TPA0122
2-W STEREO AUDIO POWER. AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SlOS247B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics In SE Mode

RL

Cc

Lowest Fntquency

30

330I1F

161 Hz

40

330l1F

120Hz
60Hz

SO

330I1F

320

330l1F

15 Hz

10,0000

330I1F

0.05 Hz

47,0000

330I1F

0.Q1 Hz

As Table 2 indicates, most of the bass response is attenuated into a 4-n load, an 8-n load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capaCitors
L.ow-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

bridged-tied load versus single-ended mode
Figure 44 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0122 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 5).

V

(rms) =

~
212

~

2
V(rms)

Power = - RL

~TEXAS

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INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

VDD

J' ;
RL
VDD

J'!
'V;

VO(PP)

2x VO(PP)

-VO(PP)

Figure 44. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-(1 speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 45. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 J.1F to 1000 J.1F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
fc =

(6)

1

23tR LCc

For example, a 68-J.1F capacitor with an 8-(1 speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD

4dB~----~~====

Figure 45. Single-Ended Configuration and Frequency Response

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-405

TPA0122
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S247B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

single-ended operation
In SE r:node (see Figure 44 and Figure 45), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 1 VN.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The internal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 46).
100

,/

-~-

V(LRMS)

IOO(avg)

Figure 46. Voltage and Current Waveforms for BTL Amplifiers

Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

~TEXAS

3-406

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

TPA0122
2-W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

ADDI
It"ATlnM
I ...
I'W'." IMa::nCIlIIATlnN
•••• _
••••• .- .... _ _ . . .

,....

...,~I

P
Efficiency of a BTL amplifier = ~
SUP

(7)

'DDavg =

and

~

f

it

o

V
RP sin(t) dt
L

=

1

V

:7t

it x RP [cos(t)] 0
L

=

2Vp
:7t R
L

Therefore,
P SUP

2 V DD Vp
:7t RL

substituting PL and Psup into equation 7,
Vp 2
Efficiency of a BTL amplifier

2Fii:

PL =Power devilered to load
Psup =Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL =Load resistance
Vp =Peak voltage on BTL load
'DDavg =Average current drawn from
the power supply
VDD =Power supply voltage
llBTL = Efficiency of a BTL amplifier

:7t Vp

2 V DD V P = 4 V DD
:7tRL

Where:

Therefore,

(8)

TlBTL

Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-il loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 3. Efficiency Vs Output Power in S-V 8-0 BTL Systems

t

Output Power

Efficiency

Peak Voltage

(W)

(%)

(V)

Internal Dissipation
(W)

0.25

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.47t

0.53

High peak voltages cause the THD to Increase.

A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

~TEXAS

INSTRUMENTS
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TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S247B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature the intemal
dissipated power at the average output power level must be used. From the TPA0122 data sheet, one can see
that when the TPA0122 is operating from a 5-V supply into a 3-n speaker that 4 W peaks are available.
Converting watts to dB:
P dB =

P

10Log~ = 1oLog 4 W = 6 dB
Pref

(9)

1W

Subtracting the headroom restriction to obtain the average listening level without distortion yields:

6 dB -15 dB =-9 dB (15 dB crest factor)
6 dB - 12 dB = -6 dB (12 dB crest factor)
6 dB - 9 dB =-3 dB (9 dB crest factor)
6 dB - 6 dB =0 dB (6 dB crest factor)
6 dB - 3 dB =3 dB (3 dB crest factor)
Converting dB back into watts:

Pw = 10PdB/10 x P ref

=

(10)

63 mW (18 dB crest factor)

== 125 mW (15 dB crest factor)

= 250 mW (9 dB crest factor)
= 500 mW (6 dB crest factor)
= 1000 mW (3 dB crest factor)
= 2000 mW (15 dB crest factor)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-0 system, the intemal dissipation in the TPA0122 and
maximum ambient temperatures is shown in Table 4.

Table 4. TPA0122 Power Rating, 5-V, 3-0, Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPAT10N
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE
-3°C

4

2W(3dB)

1.7

4

1000 mW (6 dB)

1.6

6°C

4

500 mW (9 dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

76°C

4

63 mW (18 dB)

0.6

96°C

~TEXAS

INSTRUMENTS
3-408

POST OFACE BOX 655303 • DAllAS. TExAs 75265

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

APPliCATiON iNFORiviATiON

crest factor and thermal considerations (continued)
Table 5. TPA0122 Power Rating, 5-V, S-n., Stereo
PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

2.5W

1250 mW (3 dB crest factor)

0.55

100°C

2.5W

1000 mW (4 dB crest factor)

0.62

94°C

2.5W

500 mW (7 dB crest factor)

0.59

9?DC

2.5W

250 mW (10 dB crest factor)

0.53

102°C

The maximum dissipated power, POmax , is reached at a much lower output power level for an 8 a load than for
a 3 a load. As a result, this simple formula for calculating Pomax may be used for an 8 a application:

P

f

Omax

2V >D
=-1{2R
L

(11 )

However, in the case of a 3 a load, the POmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 a load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to eJA:
El

JA

=

1
Derating Factor

=

_1_
0.022

= 450C/W

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given eJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0122 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
(13)

TA Max = T J Max - ElJA Po

=

150 - 45(0.6 x 2)

=

96°C (15 dB crest factor)
NOTE:

Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.
TableS 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0122 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-0 speakers dramatically increases the thermal performance by increasing amplifier
efficiency.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-409

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
SElBTL operation
The ability of the TPA0122 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0122, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When
SElBTL is held low, the amplifier is on and the TPA0122 is in the BTL mode. When SE/BTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0122 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SE/BTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 47.

20

RHPIN

23

RLiNEIN

R

MUX
ROUT+

8

21

RIN

VDD
ROUT-

16

100110
SEis'fL

15 100 110

Figure 47. TPA0122 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-knt1-kn divider pulls the SE/BTL input low. When a plug is inserted, the 1-kn
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the OUT-amplifier is
shutdown causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (Co) into the headphone jack.

~TEXAS

3-410

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SLOS247B - JUNE 1999 - REVISED MARCH 2000

AppLiCATiON iNFORiviATiON
PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is normally activated automatically, but may be selected
manually by pulling PCB ENABLE high. When the PC BEEP input is active, both of the LlNEIN and HPIN inputs
are deselected and both the left and right channels are driven in BTL mode with the signal from PC BEEP. The
gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is independent of the volume setting. When
the PC BEEP input is deselected, the amplifier will return to the previous operating mode and volume setting.
Furthermore, if the amplifier is in shutdown mode, activating PC BEEP will take the device out of shutdown and
output the PC BEEP Signal, then return the amplifier to shutdown mode.
When PCB ENABLE is held low, the amplifier will automatically switch to PC BEEP mode after detecting a valid
signal at the PC BEEP input. The preferred input signal is a square wave or pulse train with an amplitude of 1
Vpp or greater. To be a accurately detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall
times of less than 0.1 J.LS and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier
will return to its previous operating mode and volume setting.
When PCB ENABLE is held high, PC BEEP is selected and the LlNEIN and HPIN inputs are deactivated
regardless of the input signal. PCB ENABLE has an internal 100 kn pulldown resistor and will trip at
approximately Vool2.
If it is desired to ac couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
equation 14:
CpCB ;:: 211: fpCB \100 kQ)

(14)

The PC BEEP input can also be dc coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

~11

TPA0122
2·W STEREO AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS
SL0S247B-JUNE 1999- REVISED MARCH 2000

APPLICATION INFORMATION

Input MUX operation
Right
Headphone
Input
Signal

CIRHP
,0.47 IIF

----1

20

RHPIN

23

RLINEIN

R
CIRLINE
0.4711F

MUX

RightLine ~
Input
~
Signal

8

ROUT+

21

ROUT-

16

RIN

Figure 48. TPA0122 Example Input MUX Circuit
Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SElBTL operation section for a description of the headphone jack control circuit.

shutdown modes
The TPA0122 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, 100 = 150 /lAo SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.

Table 6. Shutdown and Mute Mode Functions
AMPLIFIER STATE

INPUTSt
SE/BTL

SHUTDOWN

INPUT

Low

High

Line

BTL

X

Low

X

Mute

High

High

HP

SE

t Inputs should never be left unconnected.
X

=do not care

~TEXAS

3-412

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

OUTPUT

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
- MAY 1999 - REVISED MARCH

• Compatioie With PC SS Deskiop Linti:-vut
Into 10-kO Load
• Compatible With PC 99 Portable Into 8-0
Load
• Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
• DC Volume Control From +20 dB to -40 dB
• 2-W/Ch Output Power Into 3-0 Load
• PC-Beep Input
• Depop Circuitry
• Stereo Input MUX
• Fully Differential Input
• Low Supply Current and Shutdown Current
• Surface-Mount Power Packaging
24-Pin TSSOP PowerPADTM

~wp PAC~_~GE

(TOP VIEW)

GND
PCB ENABLE
VOLUME
lOUT+
lLiNEIN
lHPIN
PVoo
RIN
lOUTLIN
BYPASS
GND

10
2

24
23

3

22

4

21
20
19
18
17
16
15
14
13

5

6
7
8
9
10
11
12

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
Voo
PVoo
ClK
ROUTSE/BTl
PC-BEEP
GND

description
The TPA0132 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-0 loads. This device minimizes the number of
external components needed, which simplifies the design and frees up board space for other features. When
driving 1 W into 8-0 speakers, the TPA0132 has less than 0.4% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
Amplifier gain is controlled by means of a dc voltage input on the VOLUME terminal. There are 31 discrete steps
covering the range of +20 dB (maximum volume setting) to -40 dB (minimum volume setting) in 2 dB steps.
When the VOLUME terminal exceeds 3.54 V, the device is muted. An internal input MUX allows two sets of
stereo inputs to the amplifier. In noteboOk applications, where internal speakers are driven as BTL and the line
outputs (often headphone drive) are required to be SE, the TPA0132 automatically switches into SE mode when
the SElBTL input is activated, and this effectively reduces the gain by 6 dB.
The TPA0132 consumes only 10 mA of supply current during normal operation. A miserly shutdown mode is
included that reduces the supply current to less than 150 ~.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TPA0132 to operate at full power into 8-0 loads at ambient temperatures of 85°C.

A

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~~':":1:=~"p.,~t!r:~,e:,=~.,:

Slandord warranty. Production processing"'" not nocessarlty Include
testing 01 all parameters.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3--413

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

functional block diagram
RHPIN

~

RLiNEIN - - - 1

M~X

'--...,.._..1

>--.------

ROUT+

>-......- 1 - - - - - - -

ROUT-

VOLUME-------------.

RIN

PC-BEEP
PCB ENABLE

--------+----\-.

-----1 PC
----IL._Be_ p--'
8_

Power
Management
SEtBTL

LHPIN

PVDD
VDD
BYPASS
SHUTDOWN

' - - - - - - - GND

[;gM~X

LLINEIN - - - 1

'---_..I

>-......- - 1 - - - - - -

LOUT+

>--.------

LOUT-

LIN - - - - - - - - - - - - \ - .

~TEXAS

INSTRUMENTS
3-414

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

AVAiLAtiLE OPTiONS

PACKAGED DEVICE
TA

TSSOpt
(PWP)

-40°C to 85°C

TPA0132PWP

t The PWP package IS available taped and reeled. To order a taped and reeled part,
add the suffix R to the part number (e.g., TPAOI32PWPR).

Terminal Functions
TERMINAL
NAME
NO.
BYPASS

11

CLK

17

GNO

1,12
13,24

110

DESCRIPTION
Tap to voltage divider for intemal mid-supply bias generator

I

If a 47-nF capacitor is attached, the TPA0132 generates an intemal clock. An extemal clock can override
the intemal clock input to this terminal.
Ground connection for circuitry. Connected to thermal pad.

LHPIN

6

I

Left channel headphone input, selected when SElBTL is held high

LIN

10

I

Common left input for fully differential input. AC ground for single-ended inputs.

LLiNEIN

5

I

Left channel line negative input, selected when SE/BTL is held low

LOUT+

4

LOUT-

9

0
0

Left channel negative output in BTL mode and high-impedance in SE mode

PCB ENABLE

2

I

If this terminal is high, the detection circu~ry for PC-BEEP is overridden and passes PC-BEEP through
the amplifier, regardless of its amplitude. If PCB ENABLE is floating or low, the amplifier continues to
operate normally.

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a> I-V (peak-ta-peak) square wave is input
to PC-BEEP or PCB ENABLE is high.

PVDD

7,18

I

Power supply for output stage

RHPIN

20

I

Right channel headphone input, selected when SE/BTL is held high

RIN

8

I

Common right input for fully differential input. AC ground for single-ended inputs.

RLiNEIN

23

I

Right channel line input, selected when SE/BTL is held low

ROUT+

21

Right channel positive output in BTL mode and positive output in SE mode

ROUT-

16

0
0

Left channel positive output in BTL mode and positive output in SE mode

Right channel negative output in BTL mode and high-impedance in SE mode

SE/BTL

15

I

Input MUX control input. When this terminal is held high, the LHPIN or RHPIN and SE output is selected.
When this terminal is held low, the LLiNEIN or RLiNEIN and BTL output are selected.

SHUTOOWN

22

I

When held low, this terminal places the entire device, except PC-BEEP detect circu~ry, in shutdown
mode.

VOD

19

I

Analog VOO input supply. This terminal needs to be isolated from PVOO to achieve highest performance.

I

VOLUME detects the dc level at the terminal and sets the gain for 31 discrete steps covering a range of
20 dB to -40 dB for dc levels of 0.15 V to 3.54. When the dc level is over 3.54 V, the device is muted.

VOLUME

3

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-415

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)*
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
:j: Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and

functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions' is not
Implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
pwp

=

DERATING FACTOR
2.7W§

TA 70°C
1.7W

21.8mW/"C

1.4W

§ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMAOO2), for more Information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage. VOO
High-level Input voltage, VIH

MIN

MAX

4.5

5.5

SElBTL

4

SHUTDOWN

2
3

SHUTDOWN

0.8

Operating free-air temperature. TA

-40

electrical characteristics at specified free-air temperature, Voo
noted)
PARAMETER

V
V

SElBTL

Low-level input voltage, VIL

UNIT

85

V
°C

=5 V, TA =25°C (unless otherwise

TEST CONDITIONS

MIN

TYP

MAX

VI =0, AV=2

Power supply rejection ratio

VOO=4Vt05V

IIIHI

High-level input current

VOO=5.5V,
VI=VOO

900

nA

IIILI

Low-level Input currant

VOO=5.5V,
VI=OV

900

nA

ZI

Input impedance

IVool
PSRR

100

Supply current

100(SO)

Supply current, shutdown mode

67

mV
dB

See Figure 28
BTL mode

10

15

SEmode

5

7.5

150

300

~TEXAS

3-416

25

UNIT

Output offset voltage (measured differentially)

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

mA

I1A

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

operating characteristics, Voo =5 V, TA =25"(;, RL =4 LA, Gain =:2 Y,V, BTL mode (unie$$ uiner-wise
noted)
PARAMETER

TEST CONDITIONS

MIN

TYP

Po

Output power

THO=I%,

1=1 kHz

THO+N

Total harmonic distortion plus noise

PO=1 W,

1=20Hztol5kHz

BaM

Maximum output power bandwidth

THO=5%

Supply ripple rejection ratio

1= 1 kHz,
CB=0.47 I1F

BTL mode

65

SEmode

60

CB=0.47 I1F,
1= 20 Hz to 20 kHz

BTL mode

34

Noise output voltage

SEmode

44

Vn

MAX

UNIT

2

W

0.4%
kHz

>15

dB

I1V RMS

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power
vsGain

1,4,6,8,10
2

THD+N

Total harmonic distortion plus noise

vs Output voltage

12

Vn

Output noise voltage

vs Frequency

13

SNR

VB Frequency

Supply ripple rejection ratio

vs Frequency

14,15

Crosstalk

vs Frequency

16,17,18

Shutdown attenuation

VB Frequency

19

Signal-ta-noise ratio

VB Frequency

20
21,22

Closed loop response
Po

3,5,7,9,11

Output power

Po

Power dissipation

ZI

Input impedance

vs Load reSistance

23,24

vs Output power

25,26

vs Ambient temperature

27

vsGain

28

~TEXAS

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eox 655303 •

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3-417

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL

SLOS223B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

GAIN

10%

1%

••c:+
z

1

~

1%

~

is

.!:!
c:

=
-

L

I

I

RL=8('!

RL=3('!

0

=
-

~

z

Ay = +20 toO dB
f=1 kHz
BTL

~

0.01%
0.5 0.75 1

1.25 1.5 1.75 2

2.25 2.5 2.75

I\.

"-...
0.1%

-

-

~

0.01%
-40

3

-30

Figure 1

vs

FREQUENCY

OUTPUT POWER

••zc:+

+

IS

~0

1%

1i
Q

PO=1W

u

0

PO=0.5W

!

~

0.1 O.j

r""'-r-1%

.!:!
c:

III

I

r- t--..

Ii

~~

J:

!

'7

~

0.1%

r-..

I

~

pl\ =1.75W ' ?IIIIIIII-

~

f=20kHz

IV'

~~

+

RL=3('!
Ay = +20 to OdB
BTL

Q

J:

111111
100

1k

10k 20k

0.01%
0.01

f - Frequency - Hz

Figure 3

"'

0.1
Po - Output Power - W

Figure 4

~lEXAS

INSTRUMENTS
3-418

./

f=20Hz

Z

I-

0.01%
20

20

10%

RL=3('!
Ay=+20toOdB
BTL

c

10

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10%

j

-20
-10
o
Ay • Yoltage Gain· dB

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

;:

'\

I\.

Po - Output Power - W

I

--

'z7
o

I

0

BTL

.S!

I!

//

1

0.1%

I

l-

IS

)'

I

t- Po = 1 Wfor Ay~B
~ Yo = 1 YRMS for Ays4 dB
t- RL=8('!

+

II

RL=4('! /

I

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

10

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL

---_.__

SLOS223B - MAY 1999 - REVISED MARCH 2000

-I ..yt"I\"iAL
_-- -. \"nAnA\".
-_ ... _.. enl;:' ••......,,;:,"
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

.~
+

c

~

~

+

c
0

'E0

1%

If= 20 kHz

............ ~

u

-"

..§

PO=0.25W

::t:

~

1%

'Iii
2i

~0

:e

RL=40
AV = +20 to 0 dB
BTL

j!
z0

RL=40
Av = +20 to 0 dB
BTL

z

0.1%

. / ...... ~

~F

I

Z

..§

r- ~

::t:

'ii 0.1%
PO=1.5 W

..

f=1kHz
I-..

~I

Z

t-

+

c0

+

Q

f=2OHz

Q

i=

::t:

Po=1W

I-

I I I lUll

0.01%

100
1k
f - Frequency - Hz

20

IIII

0.01%
0.01

10k 20k

0.1
Po - Output Power - W

Figure 5

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

1=
J--

.~

J-J--

z

+

vs

FREQUENCY

OUTPUT POWER
10%

RL=SO
AV = +20 to 0 dB
BTL

+

1%

.e

1%

.!!
Q

'"

Po = 0.25 W

0

~

RL=SO
AV=+20toOdB
r- BTL

r

c

~

'2

I:e

r1=

j!
z0

0

~u

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

c

'E0

o.1%

~

I

Z
+
Q
::t:

PO=0.5W

...

I-

0.01%
20

10

~

I

II

~ t===

IJl~~lkHz I

.2
c

r-

0

Ii

::t:

:e

f=1 kHz

0.1% J-...

~
I

Z

+

Q

!"

i=

PO=1W
100

1k

10k 20k

r0.01%

f=20Hz

~

I I IIII

O.ot

f - Frequency - Hz

Figure 7

0.1
Po - Output Power - W

10

Figure 8

-!!1
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-419

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B- MAY 1999- REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQU!:NCY

TOTAL HARMONIC DISTORTION PLUS NOISE
va
OUTPUT POWER

10

10%
RL=320
Ay = +14 toOdB
SE

3l

~

+
c

i
~
I!
z

0

;:

-

I

0.1 %

r=

I;;;;;

til

Po=25mW

c

~

!

i!:
0.001 %
20

....

!z

0.1%

k f=1 kHz
r--- T

I

PO=50mW rr100

+

Q

Po=75mW

i!:

JJ II III

lJllill

-

f=2OkHz

!

I"'--

0.01

f:

.2

E

~

1"00

1%

Q

jill"'"'

o

I

+

c

%

~

{}.

!z

1k
f - Frequency - Hz

0.01%
0.01

10k 20k

f=20Hz

0.1
Po - Output Power - W

Figure 9

Figure 10

TOTAL HARMONIC DISTORTION PLUS NOISE
va
FREQUENCY

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT VOLTAGE

10%

1=

10%
RL=10kO

i= Ay = +14 to 0 dB

I

I

r- SE

+

+

j

1%

II

..

Q

J

0.1%
YO=1 YRMS

!

If
z

RL=320
Ay= +14 to 0 dB
SE

1%

0.1%

!

""'"

t-

~

0.01%

100

1k
f - Frequency - Hz

10k 20k

.~

0.001%

o

• . . • L.

0.2 0.4 0.6

0.8

PO=2OHz

11

1 1.2 1A 1.6 1.8

Yo - Output Yoltage - VRMS

Figure 11

Figure 12

~TEXAS

INSTRUMENTS
3-420

~;;

RL=10kO
Ay=+14toOdB
SE

i!:
0.001%
20

I

PO=1 kHz
•..•

~

i!:

PO=20kHz

0.01%

I

z

~

I

~&

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

2

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

160

;II:

~

SUPPLY RIPPLE REJECTION RATIO

vs

vs

FREQUENCY

FREQUENCY
0

' , , "I

V~~~'5'V

BW = 22 Hz to 22 kHz
RL=4n

140

m

'0

•

,/

100

~

J!0
5

,

60

!

::f'

...

V

20

AV=+20dB

-40

.....

-60 .....

l'v ,..

Co

if

~

-80

til

-100

fo-'

o

o

lk

100

-120

10k 20k

20

lk

100

o
m

f - Frequency - Hz

Figure 13

Figure 14

I

I

vs

FREQUENCY

FREQUENCY

'

,

CB=0.47/lF
SE

-50

r"'-""
-40

-60

.........

I

AV=+6dB

V

r"""

.-60
a::

i

a::

-80

f

-100

til

CROSSTALK

vs
-40

'RL'~ 32'n

-20 -

10k 20k

f - Frequency - Hz

SUPPLY RIPPLE REJECTION RATIO

'0

......::

AV=+6dB

Co
Co
:::I

I-- r-t-'

I

111111

t

AV=+6dB

40

I

c

1

80
~

I

t

...... 10-'

AV = +20 dB

z
0

-20

I

120

I

I

R~~8n
CB=0.47/lF
BTL

~

m

'0

~~~I~~

I

RL=8n
AV= +20 dB
BTL

-70

L
LEFT TO RIGHT

I

1

.........

-80

P

0

"'"

V

~""

-90

RIGHT TO LEFT

AV=+14dB

II

-100
-110

-120
20

lk

100

10k 20k

-120
20

lk

100

f - Frequency - Hz

f - Frequency - Hz

Figure 15

Figure 16

10k 20k

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303. DAUAS. TEXAS·75265

3-421

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B

MAY 1999- REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-40

0
PO=1W
RL=SO
AV=+6OdB
BTL

-00
-60
III

"

1e
(.)

-20

1111111
LEFT TO RIGHT

-70

I

ill

-80
-80 r-

VO=1 V~MS
RL=10kO
AV=+6dB
SE

I

V

I

~I~~~:if

J...-i-'

L

.........

-40

III

"
I

1e

~o

LEFT TO RIGHT

(.)

-80

-100

""'"

I

-110
-120

20

100

1k
f - Frequency - Hz

-120

10k 20k

20

100

Figure 17

1k
f - Frequency - Hz

SIGNAL-TO-NOISE RATIO

vs

vs

FREQUENCY

FREQUENCY

0

120
VI=1 VRMS

II"
II

-20
III

III

~

RL = 10 kO,SE

-40

ic

!

~
.I::

-80

~

"I

I

J0

~o

Z

V

RL=32n,SE

f"1"'"

rn

~

-100
-120
20

PO=1 W
RL=SO
BTL

115

I

0

~

ic

DI

105

1"'"

95

1'"

I

II:

z
rn

10k 20k

-

r-- 1-1"-

t--

90
I
85

1k
f - Frequency - Hz

AV= +20 dB

i"'----.

100

iii

II RLiSti illlll
100

110

80

o

I~YI ~ +6 IdBI
1k

Figure 20

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

"""-

IIIII
100

f - Frequency - Hz

Figure 19

3-422

10k 20k

Figure 18

SHUTDOWN ATTENUATION

"c

R~~~ TOrL~"+

-100

10k 20k

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE
30

25

I~~I~I~QI

r--

1800
II

11111

Ay = +20 dB
BTL

~~:~"

20
ID

15

'tI
I
C

'ii

IIIII

~V

V

10

1\

111111

Phase

~

900

\

CI

\

5

"

o
-5

-10
10

100

1k

10k

-900

-1800

100k

1M

f - Frequency - Hz

Figure 21

CLOSED LOOP RESPONSE
30

1800

1111

RL=8Q
Ay=+6dB
BTL

25

900

20
ID

15

'tI

I

c

~

10

'1-0

Phase
1111

1111

5

~

II

"'Gain

\"

o

"

-5

-10
10

-900

100

1k

10k

100k

-1800
1M

f - Frequency - Hz

Figure 22

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-423

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B- MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

vs

LOAD RESISTANCE

LOAD RESISTANCE

3.5
Ay

2
1.5

~I

e.'$I

10%THD+N

0

\.

I

rP

~
"
I I III

t-- t---

0.5

1000

750

~

\~

I

,p

~

500

~~

250

1%THD+N

o

o

a

16

24

40

32

56

4a

o

64

o

10%THD+N

1%TH~
-.l
a

RL - Load Resistance - 0

I

~

c

0

L

I

J
I

rP

1.2

vs
OUTPUT POWER

0.8
0.6
0.4

~

-

30

lL:

l/~

JL
ILL"
rL

~

ao

0.35
I

II
a.

--

0.2
0.15

Q

2

~

/

'L t-- r--....

[I

0.1

2.5

o
o

~

Po - Outpul Power - W

U

U

ao

""
M

f= 1 kHz
BTL
Each Channel
~

~

Po - Output Power - W

Figure 26

Figure 25

3-424

r-.... ~o

......

0.05 ~
I'

Each Channel

1.5

L
V

320

BTL
0.5

0.25

I

a.

0.2

1

0

---

1=1 kHz

o
o

0.3

c
40

~

64

0.4

ii=

~

56

POWER DISSIPATION

OUTPUT POWER

L

I

~

40

vs
1.a

1.4

~

Figure 24

POWER DISSIPATION

1.6

N

RL - Load Resistance - 0

Figure 23

ii=

=+1410 OdB

1250 I

\ 1\

'5

~

Ay
SE

1

2.5

I

J

1500

=+2010 0 dB

BTL

3

ii=

OUTPUT POWER

vs

:'I
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

V

U

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION

INPUT IMPEDANCE

vs

vs

AMBIENT TEMPERATURE

GAIN

7

6

1\

~
I

c

5

~
I

4

3

0
D.
I

Q

"-

9JA3

II
9JA1,2

2

D.

o

~4

0

~

~

"r'
I'

~

~

80

a

70

~

8C

~

~

I.Iii

50

II

"- 1\

"""'" ~

--

90

I

1\

0

1&Do

9JA1 = 45.9°CIW
9JA2 = 45.2°CIW
9JA3 = 31.2°CIW 9JA4 = 18.6°CIW

\

9JA4

\

'5
Do

.5

~

~

~

\

I

"

N

~ 1\
.........

~

\

~

"

1001~1~1~

TA - Ambient Temperature - °C

30

10
-40

-30

Figure 27

-20

-10

o

10

'\
20

AV-Gain-dB

Figure 28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-425

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223S - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 1. DC Volume Control
VOLUME (Terminal 3)

GAIN of AMPLIFIER

FROM

TO

(V)

(V)

0
0.15

0.28

20
18

0.28

0.39

16

0.39

0.5

14

0.5

0.61

12

0.61

0.73

10

0.73

0.84

8

0.84

0.95

6

0.95

1.06

4

1.06

1.17

2

1.17

1.28

1.28

1.39

0
-2

1.39

1.5

-4

1.5

1.62

-6

1.62

1.73

-8

1.73
1.84

1.84
1.95

-10
-12

0.15

(dB)

1.95

2.07

-14

2.07

2.18

-16

2.18

2.29

-18

2.29
2.41

2.41
2.52

-20
-22

2.52

2.63

-24

2.63

2.74

-26

2.74

2.86

-28

2.86

2.97

-30

2.97

3.08

-32

3.08

3.2

3.2

3.31

-34
-36

3.31

3.42

3.42

3.54

-38
-40

3.54

5

-85

selection of components
Figure 29 and Figure 30 are a schematic diagrams of typical notebook computer application circuits.

~TEXAS

INSTRUMENTS
3-426

POST OFFICE BOX 655303 • DAU.AS. TEXAS 75265

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SlOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Right CIRHP
Head- 0.47 j1F
phone
Input
Signal
20

-1

RHPIN

R
CIRLINE
Right 0.47 j1F
LIne
Input
Signal

23

RLINEIN

MUX

-1

8

ROUT+

21

ROUT-

16

RIN

CRIN
0.47 jLF

T
-=PC BEEP
14
Input
Signal CPCB
0.47 11F 2
VDD

--1

r

100kn

-=VOLUME
ClK

w

SElBTl

CClK
-=-47nFT

Gain!
MUX
Control

PVDD
Depop
Circuitry
Power
Management

Left CllHP
Head- 0.47j1F
phone
Input
Signal

-1

6

18 See Note A

VDD
CSR

VDD

19

BYPASS
SHUTDOWN

11

lHPlN

-:J' 0.1j1F

VDD

T

22

GND

CSR
0.1j1F
CBYP

To

-:J' 0.47 j1F

System-

CllLlNE
left 0.47 j1F
LIne
Input
Signal

-1

lOUT+

Control
1,12
4
13,24

lOUT-

9

1 kn

LIN
CLIN
OA7j1F

-=-

100kn
NOTE A.

A 0.1 j1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals. a larger
electrolytic capacitor of 10 j1F or greater should be placed near the audio power amplifier.

Figure 29. Typical TPA0132 Application Circuit Using Single-Ended Inputs and Input MUX

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-427

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
N/C
20
CCRINRight 0.47 ILF
23
Negativ;H
Differential
Input
Signal

CRIN+
Right 0.47 1LF
8
Positive
Differential
Input
Signal
PC BEEP
14
Input
Signal Cp B
0.471LF 2

-1

ROUT+

21

RIN

CoUTR
330ILF

--::1

PCB
ENABLE

ROUT-

16

VDD

Beep

VDD

1 kQ

100kQ

r~

VOLUME
CLK
SE!BTL

CCLK
-::- 47nFT

Gain!
MUX
Control

Depop
Circuitry

'J:'

Power
Management

CIIHP
Head- 0.47 ILF

Left

-'-1

phone
Input
Signal

6

CILUNE

5

LHPIN
LUNEIN

L
MUX

L

_--A.iV\r____

PVDD
18 See Note A
I---'-'='-f---'-"-----il...-- VDD
CSR
0.11LF
VDD 19
VDD
BYPASS
SHUTDOWN

~w\:==;-rJ--G=.:..:NDIl

11 '
22

--1

CSR
0.11LF
CBYP

'J:' 0.471LF

To
SystemControl

Left 0.471LF
Line
Input
Signal

T

LOUT+

4

LOUT-

9

1 kQ

1,12,
13,24

CoUTR
330ILF

UN
CUN
0.471LF T

-::-

100kQ
NOTE A.

A 0.1 ILF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 ILF or greater should be placed near the audio power amplifier.

Figure 30. Typical TPA0132 Application Circuit Using Differential Inputs

~TEXAS

INSTRUMENTS
3-428

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r------------

Input Signal

I
I
I

Rf

----j I--_-..:.:..:...-t-..J\j\/\r--*-I
R

Figure 31. Resistor on Input for Cut-Off Frequency
The input resistance at each gain setting is given in Figure 28.
The -3 dB frequency can be calculated using the following formula:
f

1
-3 dB - 2,,; C(R II RI)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.

input capacitor, CI
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, Z" form a
high-pass filter with the corner frequency determined in equation 2.

(2)
fC(highpaSS) =

2,,;iIN C I

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-429

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL

SLOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, CI (continued)
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where ZI is 710 k.Q and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.

C -

1

I - 2ltZ I fo

(3)

In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 ~F. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
that the source dc level. Note that it is important to confirm the capacitor polarity in the application.

power supply decoupllng, Cs
The TPA0132 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 ~F placed as close as possible to the device Voo. lead works best. For
filtering lower-frequency noise Signals, a larger aluminum electrolytiC capacitor of 10 ~F or greater placed near
the audio power amplifier is recommended.

midrail bypass capacitor, CBYP
The mid rail bypass capacitor, CSyp, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CSyp determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the mid rail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CSyp, values of 0.47 ~F to 1 ~F ceramic or tantalum low-ESR capacitors are recommended
for the best THO and noise performance.

~TEXAS

3-430

INSTRUMENTS
POST OFROE BOX 655303 • DAUAS. TEXAS 75265

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

(4)

fC(hI9h)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 !iF is chosen and loads vary from 3 0,
4 0, 8
320, 10 kil, and 47 kil. Table 2 summarizes the frequency response characteristics of each
configuration.

n.

Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode

Cc

RL

Lowest Frequency

30

330l1F

161 Hz

40

33011F

120Hz
60 Hz

ao

33Ol1F

320

33011F

15 Hz

10,0000

330l1F

0.05 Hz

47,0000

33011F

0.01 Hz

As Table 2 indicates, most of the bass response is attenuated into a 4-0 load, an 8-0 load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capaCitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-431

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load versus single-ended mode
Figure 32 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0132 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 5).
V

(5)

_ V O(PP)
(nns) 212
V(nns)

2

Power = - -

RL

VDD

V'

RL

~

J'!
rv ~

VO(PP)

2xVO(PP)

-VO(PP)

Figure 32. Bridge-Tied Load Configuration

~TEXAS

3-432

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS2238 - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-n speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 33. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 3311F to 1000 I1F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
f

-

(c) -

(6)

1
23tR L Cc

For example, a 68-I1F capacitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD

~dB~-----J~=====

Figure 33. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

single-ended operation
In SE mode (see Figure 32 and Figure 33), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 1 VN.
.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The internal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the internal
power dissipation of the amplifier.

-!!1

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-433

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 34).
IDD

,/

-~-

V(LRMS)

IDD(avg)

Figure 34. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
P

'Efficiency of a BTL amplifier = ~
SUP

and

(7)

looavg

=~

fo'"

V
RP sin(t) dt

=~ x

L

V
It
RP [oos(t)] 0
L

= 2Vp
It

RL

Therefore,
_ 2 Voo Vp
PSUP It RL
substituting PL and Psup into equation 7,
Vp2

Efficiency of a BTL amplifier
Where:

2F\
2Voo Vp
It RL

Therefore,

PL =Power delivered to load
PSUP =Power drawn from power supply
VLRMS =RMS voltage on BTL load
RL =Load resistance
Vp =Peak voltage on BTL load
looavg =Average current drawn from
the power supply
Voo = Power supply voltage
TlBTL =Efficiency of a BTL amplifier
(8)

-!!1 TEXAS

3-434

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 3. Efficiency Vs Output Power in 5-V 8-0 BTL Systems
Output Power

Efficiency
(%)

Peak Voltage
(V)

Internal Dissipation

(W)

0.25

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.47t

0.53

(W)

t High peak voltages cause the THD to increase.
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature the internal
dissipated power at the average output power level must be used. From the TPA0132 data sheet, one can see
that when the TPA0132 is operating from a 5-V supply into a 3-0 speaker that 4 W peaks are available.
Converting watts to dB:
P dB

=

10Log Pw
P ref

=

10Log 41 Ww

=

6 dB

(9)

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6
6
6
6
6

dB -15 dB = -9 dB (15 dB crest factor)
dB - 12 dB = -6 dB (12 dB crest factor)
dB - 9 dB -3 dB (9 dB crest factor)
dB - 6 dB 0 dB (6 dB crest factor)
dB - 3 dB 3 dB (3 dB crest factor)

=
=
=

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

3-435

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Converting dB back into watts:
Pw = 10PdB/10 x P ref

(10)

= 63 mW (18 dB crest factor)
= 125 mW (15 dB crest factor)

=
=

250 mW (9 dB crest factor)
500 mW (6 dB crest factor)

= 1000 mW (3 dB crest factor)

=

2000 mW (15 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-n system, the internal dissipation in the TPA0132 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0132 Power Rating, 5-V, 3-n, Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE
-3°C

4

2W(3dB)

1.7

4

1000 mW (6 dB)

1.6

6°C

4

500 mW (9 dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

78°C

4

63mW(18dB)

0.6

96°C

Table 5. TPA0132 Power Rating, 5-V, a-n, Stereo
(W/Chennel)

MAXIMUM AMBIENT
TEMPERATURE

1250 mW (3 dB crest factor)

0.55

100°C

1000 mW (4 dB crest factor)

0.62

94°C

2.5W

500 mW (7 dB crest factor)

0.59

97°C

2.5W

250 mW (10 dB crest factor)

0.53

102°C

PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

2.5W
2.5W

POWER DISSIPATION

The maximum dissipated power, POmax' is reached at a much lower output power level for an 8 n load than for
a 3 n load. As a result, this simple formula for calculating POmax may be used for an 8 n application:

P
Omax

= 2Vf>D
n2R
L

(11)

However, in the case of a 3 n load, the POmax occurs at a point well above the normal operating power level.
The amplifier may tl:lerefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 n load.

~TEXAS

.

INSTRUMENTS

POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to 9JA:
e

JA

=

1
Derating Factor

= _1_
0.022

= 450C/W

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given 9JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0132 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
TA Max = TJ Max -

e JA

(13)

Po

= 150 - 45(0.6 x 2) = 96°C (15 dB crest factor)
NOTE:

Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.

Tables 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0132 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Table 4 and 5 were calculated for maximum listening
volume without distortion. When the output level is reduced the numbers in the table change significantly. Also,
using 8-0 speakers dramatically increases the thermal performance by increasing amplifier efficiency. '

SE/BTL operation
The ability of the TPA0132 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0132, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT- and ROUT- (terminals 9 and 16). When
SElBTL is held low, the amplifier is on and the TPA0132 is in the BTL mode. When SElBTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0132 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SElBTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 35.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-437

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SlOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

20

RHPIN

23

RLiNEIN

R

MUX
ROUT+

8

21

RIN

VDD
ROUT-

16

100kn
SElBfi:

15r1_00_kn
__

p

Figure 35. TPA0132 Resistor Divider Network Circuit

Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 100-kn/1-kO divider pulls the SEfBTL input low. When a plug is inserted, the 1-kO
resistor is disconnected and the SEfBTL input is pulled high. When the input goes high, the OUT-amplifier is
shutdown causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (Co) into the headphone jack.

PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is normally activated .activated automatically, but may be
selected manually by pulling PCB ENABLE high. When the PC BEEP input is active, both of the LlNEIN and
HPIN inputs are deselected and both the left and right channels are driven in BTL mode with the signal from
PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is independent of the volume
setting. When the PC BEEP input is deselected, the amplifier will return to the previous operating mode and
volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC BEEP will take the device out
of shutdown and output the PC BEEP signal; then return the amplifier to shutdown mode.
When PCB ENABLE is held low, the amplifier will automatically switch to PC BEEP mode after detecting a valid
signal at the PC BEEP input. The preferred input signal is a square wave or pulse train with an amplitude of 1
Vpp or greater. To be a accurately detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall
times of less than 0.1 IJ.S and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier
will return to its previous operating mode and volume setting.

~TEXAS

- INSTRUMENTS
3-438

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0132
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
When PCB ENABLE is held high. PC BEEP is selected and the LlNEIN and HPIN inputs are deactivated
regardless of the input signal. PCB ENABLE has an internal 100 kn pulldown resistor and will trip at
approximately Vool2.
If it is desired to ac couple the PC BEEP input. the value of the coupling capacitor should be chosen to satisfy
the following equation:
CpCB ;;,; 211: fpCB 1(100 kQ)

(14)

The PC BEEP input can also be dc coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.

Input MUX operation
Right
Headphone
Input
Signal

CIRHP
0.47 I!F

----1

20

RHPIN

R
CIRLINE
0.471!F
RlghtLine ~
Input
~
Signal

"'--:"+=="-1

8

MUX
ROUT+

21

ROUT-

16

RIN

T
Figure 36. TPA0132 Example Input MUX Circuit

Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SElBTL operation section for a description of the headphone jack control circuit.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-439

TPA0132
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS223B - MAY 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

shutdown modes
The TPA0132 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, Ipp = 150 ~. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.
Table 6. Shutdown and Mute Mode Functions
AMPLIFIER STATE

INPUTSt
SE/BTL

SHUTDOWN

INPUT

Low

High

Line

BTL

X

Low

X

Mute

High

High

HP

SE

t

Inputs should never be left unconnected.
X = do not care

~TEXAS

INSTRUMENTS
3-440

POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

OUTPUT

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
• Compatible With PC 99 Desktop Line-Out
Into 10-kQ Load
• Compatible With PC 99 Portable Into 8-Q
Load
• Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
• DC Volume Control From 20 dB to -40 dB
• 2-W/Ch Output Power Into 3-Q Load
• PC-Beep Input
• Depop Circuitry
• Stereo Input MUX
• Fully Differential Input
• Low Supply Current and Shutdown Current
• Surface-Mount Power Packaging
24-Pin TSSOP PowerPADTM

PWPPACKAGc
(TOP VIEW)

GND
PCB ENABLE
VOLUME
lOUT+
lLiNEIN
lHPIN
PVoo
RIN
LOUTLIN
BYPASS
GND

10
2
3
4
5
6
7
8
9
10
11
12

24
23
22
21
20
19
18
17
16
15
14
13

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
Voo
PVoo
ClK
ROUTSElBTl
PC-BEEP
GND

description
The TPA0142 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-Q loads. This device minimizes the number of
external components needed, which simplifies the design and frees up board space for other features. When
driving 1 Winto 8-Q speakers, the TPA0142 has less than 0.22% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
Amplifier gain is controlled by a dc voltage input on the VOLUME terminal. There are 31 discrete steps covering
the range of 20 dB (maximum volume setting) to -40 dB (minimum volume setting) in 2 dB steps. When the
VOLUME terminal exceeds 3.54 V, the device is muted. An internal input MUX allows two sets of stereo inputs
to the amplifier. In notebook applications, where internal speakers are driven as BTL and the line outputs (often
headphone drive) are required to be SE, the TPA0142 automatically switches into SE mode when the SElBTL
input is activated, and this effectively reduces the gain by 6 dB.
The TPA0142 consumes only 20 rnA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to less than 150 IJA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°CIW are truly realized in multilayer PCB
applications. This allows the TPA0142 to operate at full power into 8-Q loads at ambient temperatures of 85°C .

.A.

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
,

PowerPAD is a trademark of Texas Instruments Incorporated.

~~""::::'::=..I;'~r::"'~:,c=.::

slandard warranty. Pro~UCllon processing does not n_1V Includo
tooting of all poramotors.

~TEXAS

INSTRUMENTS
POST OFFiCE BOX 655303 • DALLAS. TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3-441

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
W1TH DC VOLUME CONTROL
SL0S248B - JUNE 1999 - REVISED MARCH 2000

functional block diagram
RHPIN
RLiNEIN _ _ _

~

M~X

'"--r-......

>--+-------

ROUT+

>--+-~-----

ROUT-

VOLUME-------.

RIN

PC-BEEP
PCB ENABLE

- - - - - - - - + - - - - 1 -..

----1 PC
----i.._B_ee_p.....
Power
Management

SElBTL

LHPIN

' - - - - - GND

~

LLiNEIN - - - - i

LIN

PVOD
VDD
BYPASS
SHUTDOWN

M~X

' - - - _.....

>--+--+------

LOUT+

>--+-------

LOUT-

------------11--.

~1ExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

AVAILAtlLi: OPTiONS
PACKAGED DEVICE
TA

TSSOpt
(PWP)

-40°C to 85°C

TPA0142PWP

t The PWP package IS available taped and reeled. To order a taped and reeled part,
add the suffix R to the part number (e.g., TPAOI42PWPR).

Terminal Functions
TERMINAL
NAME
NO.
BYPASS

11

CLK

17

GND

1,12
13,24

110

DESCRIPTION
Tap to voltage divider for internal mid-supply bias generator

I

If a 47-nF capacitor is attached, the TPA0142 generates an internal clock. An external clock can override
the intemal clock input to this terminal.
Ground connection for circuitry. Connected to thermal pad

LHPIN

6

I

LIN

10

I

Common left input for fully differential input. AC ground for single-ended inputs

LLiNEIN

5

I

Left channel line negative input, selected when SEIBTL is held low

LOUT+

4

0

Left channel positive output in BTL mode and positive output in SE mode

LOUT-

,9

0

Left channel negative output in BTL mode and high-impedance in SE mode

PCB ENABLE

2

I

If this terminal is high, the detection circuitry for PC-BEEP is overridden and passes PC-BEEP through the
amplifier, regardless of its amplitude. If PCB ENABLE is floating or low, the amplifier continues to operate
normally.

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > I-V (peak-to-peak) square wave is input to
PC-BEEP or PCB ENABLE is high.

7, 18

I

Power supply for output stage

20

I

Right channel headphone input, selected when SElBTL is held high

RIN

8

I

Common right input for fully differential input. AC ground for singl&'ended inputs

RLiNEIN

23

I

Right channel line input, selected when SE/BTL is held low .

ROUT+

21

0

ROUT-

16

0

Right channel negative output in BTL mode and high-impedance in SE mode

I

Input MUX control input. When this terminal is held high, the LHPIN or RHPIN and SE output is selected.
When this terminal is held low, the LLiNEIN or RLiNEIN and BTL output are selected.

PVDD
RHPIN

SElBTL

15

Left channel headphone input, selected when SE/BTL is held high

Right channel positiVe output in BTL mode and positive output in SE mode

SHUTDOWN

22

I

When held low, this terminal places the entire device, except PC-BEEP detect circuitry, in shutdown mode.

VDD

19

I

Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance.

VOLUME

3

I

VOLUME detects the dc level at the terminal and sets the gain for 31 discrete steps covering a range of
20 dB to -40 dB for de levels of 0.15 V to 3.54. When the dc level is over 3.54 V, the device is muted.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-443

TPA0142
.
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, V, ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40 c C to 85c C
Operating junction temperature range, TJ .......................................... -40c C to 150c C
Storage temperature range, Tstg .................................................. -65c C to 150°C
Lead temperature 1,.6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

PWP

DERATING FACTOR

2.7wt

21.8mW/"C

=

TA 70°C
1.7W

1.4W

:j: Please see .the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

(literature number SLMA002), for more Information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
High-level input voltage, VIH

MIN

MAX

4.5

5.5

SElBTL

4

SHUTDOWN

2

SElBTL

Low-level input voltage, VIL

0.8
:"40

Operating free-air temperature, TA

V
V

3

SHUTOOWN

UNIT

85

V
°C

electrical characteristics at specified free-air temperature, Voo = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONOmONS

Output offset voltage (measured differentially)

VI=O, AV=2

Supply ripple rejection ratio

VOO=4.9Vt05.1 V

IIIHI

High-level input current

VOO = 5.5 V, VI = VOO

IIILI

LOW-level input current

VOO=5.5V, VI=OV

IVosl

100

Supply current

IOO(SO)

Supply current, shutdown mode

MIN

TYP

20

SEmode

10
150

~TEXAS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

UNIT

25

mV

900

nA

900

nA

dB

67

BTL mode

INSTRUMENTS

MAX

mA
300

jJ.A

TPA0142
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S248B - JUNE 1999 - REVISED MARCH 2000

operating characteristics, VOO
noted)

=5 V, TA =25 C, Hl =4 fl., Gain =2 VN, aTL mode (Uiileiiii utherwise
u

PARAMETER

TEST CONDITIONS

Po

Output power

THO = 1%,

f= 1 kHz

THO+N

Total harmonic distortion plus noise

PO=lW,

f=20Hzto15kHz

BOM

Maximum output power bandwidth

THO = 5%

Vn

MIN

TYP

MAX

UNIT

2

W

0.22%
kHz

>15

Supply ripple rejection ratio

f= 1 kHz, CB = 0.4711F

Noise output voltage

CB=0.47 I1F,
f= 20 Hz to 20 kHz

BTL mode

65

SEmode

60

BTL mode

34

SEmode

44

dB

I1V RMS

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power
THO+N

Total harmonic distortion plus noise

Vn

Output noise voltage

SNR

vsGain
vs Frequency

2
3,5,7,9,11

vs Output voltage

12

vs Bandwidth

13

Supply ripple rejection ratio

vs Frequency

14,15

Crosstalk

vs Frequency

16,17,18

Shutdown attenuation

vs Frequency

19

Signal-ta-noise ratio

vs Bandwidth

20
21,22

Closed loop respone
Po

1,4,6,8,10

Output power

Po

Power dissipation

ZI

Input impedence

vs Load resistance

23,24

Output power

25,26

VS

vs Ambient temperature

27

vsGain

28

~TEXAS

INSTRUMENTS
" POST OFFlCE BOX 655303 • DAllAS, TEXAS 75265

3-445

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL

SLOS248B -JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

va

va

OUTPUT POWER

VOLTAGE GAIN

10%

1%

Iz

1

~

L

1%

f=
r--

~

!.
::c
j

I

I

+
c

I

I

RL=4D.!

I

I
If

RL=8D.

t- Po = 1 W for AV~B
~ VO= 1 VRMS for AyS4 dB
t- RL=8D.

I

-

+

RL=3D.

=
-

I I

IS

BTL

\

.2

0.1%

I

I :j

0.1%

I

'".......

j

~

7

I

Z

~

::c

AV = +20 to 4 dB
f= 1 kHz
BTL

I-

t-

0.01%
0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 2.75

3

I'..
0.01%
-40

-30

Po - Output Power - W

Figure 1

vs

FREQUENCY

OUTPUT POWER
RL=3D.
AV = +20 to +4 dB
BTL

CD

~

Z

+

+
c
0

;:

1%
PO=0.5W ~

.2

j

20

10%

RL=3 D.
AV= +20 to 0 dB
BTL

IS

I

10

TOTAL HARMONIC DISTORTION PL.US NOISE

vs
10%

I

-20
-10
o
AV· Voltege Gain· dB

~

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

I

-..........

~

-

"'........

PO=1W
0.1

..,

~

.~
0

l"""- I"'""-

.....

~
::c

~

j 0.1% F= F
~I

7
~

I

1%

f=2OkHz

1=1

k~1

f=~"'"

I""

V

Z

+

PO=1.75W -

~
0.01%
20

100

1k
f - Frequency - Hz

II '"''10k

Q

~
20k

0.01%
0.01

Figure 3

0.1
Po - Output Power - W

Figure 4

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265,

10

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

100/0

vs

FREQUENCY

OUTPUT POWER
100/0

.II
+

+

~

10/0

i

RL=40
AV = +20 to +4 dB
BTL

I

RL=40
AV = +20 to +4 dB
BTL

~

c
0
'E

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

10/0

!

f=2OkHz

is

~

~

..~
:z:

.".-

i

~

~

PO= 0.25 W

0.10/0

i

I

Z
+
Q

Po=1.5W -

j:
0.010/0
20

rrWil

I II ""
111111

100

1k
f - Frequency - Hz

I

10k 20k

~

I

1000

0.10/0

.....

I"-~

r - l - f-

~

f=1kHz

blow

f=20Hz

j:
0.010/0
0.01

0.1
Po - Output Power - W

Figure 5

81

~

t::
r-

I+

10

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

100/0

rv

I

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

1=
100/0~~~:=1.m.

RL=80
Av = +20 to +4 dB
BTL

RL=80

t- AV = +20 to +4 dB
t- BTL

c
0

1:

10/0

S

is

.2
c
PO=O.25W

@

:!
i

0.10/0

~
I

E
r-

PO=0.5W

,.

.....

Z

~

~

j:
PO=1W

0.010/0
20

100

1k

10k 20k

f - Frequency - Hz

Po - Output Power - W

Figure 7

Figure 8

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-447

TPA0142
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS2488 - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%
RL=320
AV = +14 to +4 dB
SE

I

+
o
c

+

c

1%

~

I

I

~

0.1%

r-~

~~O :~~'U

~

~

1%

ic

~
~

i=

I 11:;~t;1

0.001%
20

100

~ f=20kHz

.~
0

IS

.-"! r;II"

I::::?'

b

0.01%

z

RL=320
AV=+14to+4dB
SE

·z1

r--I-o

0.1%

{!.
I

t-

~kHZ

Z
+
C

:c

~<;> =75

..... 1===1:::

I-

III1111
1k
f - Frequency - Hz

0.01%
0.01

10k 20k

f=20Hz
0.1
Po - Output Power - W

Figure 9

Figure 10

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT VOLTAGE

10%

1=

I

10%
RL=10kO

~ AV=+14toOdB

I- SE

L,

VI

+

5

1%

I..

J

0.1%

~

z

0.1%

~

0.01%

IS

VO=1 VRMS

~
0.01%

r-

z+

~

,r-- r~~

.~
o

i=
0.001%
20

100

1k

10k 20k

f=11kHZ

v

...

"

f=20Hz

I I

0.001%
0.2 0.4

f - Frequency - Hz

0.6

0.8

1.2

1.4

1.6

Vo - Output Voltage - VRMS

Figure 11

Figure 12

~TEXAS

INSTRUMENTS
3-448

/

RL=10kO
Av=+14to+4dB
SE

c

i!=

/
f=20kHz

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

1.8

2

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

...., ........ A. ....U A CO A I"TII:COIC!TII"C!
I I

rnl,"""" ""1""',"""" I ......... I

OUTPUT NOISE VOLTAGE

SUPPLY RIPPLE REJECTION RATIO

vs

vs

BANDWIDTH

FREQUENCY

160

!!

a:
~

0

1/
100

~
III

-5z

80

:i

60

I

40

t
0

/
Ay= +20 dB

,/"

V

,f
20

III

-20

ia:

-40

'0
I
0

120

I

i

R~~8Q

YOO=5Y
RL=4Q

140

ta:

'il'

III~V

:::I

III I

"I'"I'

A7+2OdB

V

\

Ay=+6dB

-100

;1111
100

-120

10k 20k

1k
BW - Bandwidth - Hz

CB = 0.47 I1F
BTL

~
atil

II IIIII
100

-60

t

Ay=+6 dB

1--"1""

c

,e.
a: -60

~

10-'"

o!o

.~_

20

Figure 13

1k
f - Frequency - Hz

Figure 14

SUPPLY RIPPLE REJECTION RATIO

0

III

'0
I

ia:
ta-

ir

vs

FREQUENCY

FREQUENCY

I

-40

-60 t-

"'" ~T'"

-60

til

Ay=OdB

\
.....

-60

i

:::I

-50

.... 1'

~

~\

RL=8Q
AV= +20 dB
BTL

-70

...:

-80 ~~

(J

-90

Ie

1-""""

Ay=+14dB

LEFT TO RIGHT , / j.;"

RIGHT TO LEFT

-100

-100

L

III I

III
'0
I

I

~~~11IW

I

CB =0.47I1F
-20 I- SE

c

tl

CROSSTALK

vs
-40

I~L'~32'Q

10k 20k

r- I-"f-"

"""

II

-110

-120
20

100

1k
f - Frequency - Hz

10k 20k

-120
20

100

1k

10k 20k

f - Frequency - Hz

Figure 15

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-449

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S248B - JUNE 1999 - REVISED MARCI-I2000

TYPICAL CHARACTERISTICS
CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-40

PO=1W
RL=80
Ay = +6 dB
BTL

-50

-60
ID

'g

IIllll

-70

I

I

~

I
0

f8

t~~TOIRIGHT

-80

-so r-

YO=1 YRMS
RL=10kn
Ay=+6dB
SE

-50

.....

II

11111

-70 H-H+t+IfI-+-+++ttttt--+-+-t-Htttt---l

I

RIGHT TO LEFT

:!!!

-80 \ooooI;::;;H+tHIl--+-++ LEFT TO RIGHT

J -sor~~~~~~1~1I~~~~'~~~4

-100

I'
1"
RIGHT TO LEFT
-100 1-+-H+tHIl--+-'---'-TTT"fl"r-----,--+-t-H+ttt---l

-110

-110 H+t-Htlt-+-+-+t-t+tti--t-+-H-ttttt---t

-120
20

100

1k

-120 L-L...L..I..J..U.I.L-..L-.l-L.J..U.w.....--'--'-L...L.L~:--:-'.
20
100
1k
10k 20k
f - Frequency - Hz

10k 20k

f - Frequency - Hz

Figure 17

Figure 18

SHUTDOWN ATTENUATION

SIGNAL·fo-NOISE RATIO

vs

vs

FREQUENCY

BANDWIDTH

0

120
YI=1 YRMS

IIII

-20
ID

'g

0

ID
'g

RL=10kn,SE

I---

-40

!c

~

-80

11
::s

-80

i

I

I

i=
~

RL=32n,SE

.c
II)

~

-120
20

iiic
Q

110

r....

r-..... r--

105

'r--

100
95

~

~

II:

r- ""'t.....

..... r-

z

90

~

r-..

II)

85

1k

10k 20k

80

o

f - Frequency - Hz

Figure 19

100
1k
BW - Bandwidth - Hz

Figure 20

~TEXAS

INSTRUMENTS
3-450

=+20 dB

I

Rrrfljn~
100

--"

Ay

Ay=+6 dB

iii

t"-r--100

PO=1W
RL=80
BTL

115

I

c

r--

POST OFFICE BOX 855303 • DAllAS. TEXAS 75265

10k 20k

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS

CLOSED LOOP RESPONSE

30

1~~I~I~OI

25 _ AV=+2OdB

~~:~II

BTL

goo

20
15
III

"
I

~

180°

II

III

")
I

10

II

~~~~

....

',,-

5

o

-90°

-10
10

_180°

100

1k

10k

100k

1M

f - Frequency - Hz

Figure 21

CLOSED LOOP RESPONSE

30

180°

RL~8'n
AV=+6dB
BTL

25

goo

20
III

15

~

10

"cI

r--..

Phase

"'

0°

I

1111

5

~
\~

Gain

o

-900

1\

-5

-10
10

_180°

100

1k

10k
f - Frequency - Hz

100k

1M

Figure 22

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-451

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

vs
LOAD RESISTANCE

vs
LOAD RESISTANCE

3.5
3
~

1500

Ay = +20 toOdB
BTL

Ay = +14toO dB
SE

\

1250

2.5

I

I

2

.&

1.5

\
10%THD+N

~\
\~

'S

8I
rP

o

1000

I

750

'S

~

0

[).:

f\ ~ t--

0.5

~I

~

J:>
t....

~

500

\

250

1%TH~

1%THD+N

IIIII

o

8

~

~

~

~

~

o

M

$

~ 10%THD+N

o

I

I

8

~

RL - Load Resistance - 0

Figure 23

~

1.4

c:

/
//
II

I

1.2

I

0.6

POWER DISSIPATION
OUTPUT POWER

0.4

.."..-

40

-

30

~

---

I

0.3

i

0.25

Q

0.2

,/

c:

J
0

Q.

--

r--..

.......

i'-

1

1/ ~ J'-...

0.15

rl

2

80

"-l'

Q

1.5
Po - Output Power - W

r--.!.0

1

I
Q.

0.1
320

f= 1 kHz
BTL
Each Channel

0.5

M

0.4

I

0.2

o
o

$

vs

0.35

80

V

~

vs
OUTPUT POWER

V

I V-'
rP
0.8

~

POWER DISSIPATION

~

I

~

Figure 24

1.8
1.6

~

RL - Load Resistance - 0

0.05
2.5

If--..J

o
o

~

'=1 kHz
BTL
Each Channel

U

U

M

~

~

Po - Output Power - W

Figure 25

Figure 26

~TEXAS·
INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

~

~

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S24BB - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION

vs
AMBIENT TEMPERATURE
7

I

\

9JA4

6

==I
c

\

5

0

I
I
I

rP

4

"-

jJA3

1

3
9JA1,2

.1. . 1

=45.9°CIW
=45.2°C/W
=31.2°CIW
=18.6°CIW

_

\

1\
".......

"- ~ \

2

o

I

9JA1
9JA2
SJA3
9JA4

~

"\
~~
........

~~

~~o

0
~
~
60 80 1001~1~160
TA - Ambient Temperature - °C

Figure 27
INPUT IMPEDANCE

vs
GAIN

90

80

a

70

I

8c

60

I.5

50

as

s

Q.

.5

-- "" '\

,

~

\

40

I

N

30

\

20
10
~

\

-30

-~

-10
AV-Galn-dB

o

10

"

20

Figure 28

-!I1TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-453

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL

SLOS2488 - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

0.47 11F
1

""_" I Of.
-

3
4
0.4711F

LLiNEIN Of---

0.47 11 F

II

LHPOf---

II

RIN~

8

II

..I

I

9

0.4711F

ri
0.47 I1F

VOLUME SHUTDOWN
LOUT+

If---i LLiNEIN
,--!-

10
11

=i=
J

12

GND

PCB ENABLE RLINEIN

6 LHPIN

0.4711F

LOUT-OI----

GND

PVDD
RIN
LOUTLIN
BYPASS
GND

ROUT+
RHPIN
VDD
PVDD
CLK
ROUTSElBTL
PC-BEEP
GND

I

~
23 -=-

0 RLINE

22

SHUTDOWN

21
20

.r-.

0.~7I1F

II

19

I

18
47nF

~8
16

=f

,-J:-

GND

-=-

,....
~

14

~
-

~TEXAS

3-454

POST OFFICE BOX 655303 • DALu\S. TEXAS 75265

SElBTL

i~
0.47 11F

Figure 29. Typical TPA0142 Application Circuit

INSTRUMENTS

1

0 RHP

-L
~ VDD
0.1I1FT10I1F~

15

13

0.111 F

ROUT-

PC-BEEP

ROUT-

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS2488 - JUNE 1999 - REVISED MARCH 2000

...... '1' ...... "' .... ''''I:'naa.
ATlnt.1
...,,1 .... ,....
I~'.

MrrL.I"'" I 1"'1'" ......

Table 1. DC Volume Control
VOLUME (Terminal 3)

GAIN of AMPLIFIER
(dB)

FROM
(V)

TO
(V)

0
0.15

0.15
0.28

0.28

0.39

18
16

0.39

0.5

14

0.5

0.61

12

0.61
0.73

0.73

10

0.84

8

0.84

0.95

6

0.95

1.06

4

1.06

1.17

2

1.17

1.28

1.28
1.39

1.39
1.5

0
-2

1.5

1.62

1.62

1.73

-8

1.73

1.84

-10

1.84

1.95

-12

1.95

2.07

-14

2.07

2.18

-16

2.18

2.29

-18

2.29

2.41

-20

2.41

2.52

-22

2.52

2.63

-24

2.63

2.74

-26

2.74

2.86

-28

2.86

2.97

-30

2.97

3.08

-32

3.08

3.2

-34

3.2

3.31

-36

3.31

3.42

-38

3.42

3.54

-40

3.54

5

-85

20

-4
-6

selection of components
Figure 30 and Figure 31 are a schematic diagrams of typical notebook computer application circuits.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA0142
..
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Right CIRHP
Head- 0.47 J1I'
phone
Input
Signal
20

-1

CIRLINE
Right 0.4711F
Line
Input
Signal

23

RHptN

RLiNEIN

R
MUX

-1

8

ROUT+

21

ROUT-

16

RIN

CRIN
0.4711F

T
-=PC BEEP
14
Input
Signal CPCB
0.47J11' 2
VOO

---1

r~
-=-

-=-

PC-BEEP

PC-

100kn
VOLUME
ClK
SElBTl

CClK
47nFT

Galnl
MUX
COntrol

PVDD
Depop
Circuitry
Power
Management

Left

CllHP
Heed- 0.4711f
phone
Input
Signal

-1

6

CllLiNE
left 0.47 11F
Line
Input
Signal

VDD

Beep

LHPIN

VDD

19

BYPASS
SHUTDOWN

11

GND

5

-1

18

22

Sea Note A
VDD

CSR
-:;r 0.111F

VDO

T

P

CSR
0.1J11'

-=-

CBYP
-:;r 0.47
I1F

To
System
COntrol

lOUT+

4

lOUT-

9

1 kO

1,12
13,24

-=-

-=-

COUTR
33OI1F

LIN
CLiN
0.4711F

-=-

100kn
NOTE A.

A 0.1 I1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 I1F or greater should be placed near the audiO power amplifier.

Figure 30. Typical TPA0142 Application Circuit Using Single-Ended Inputs and Input MUX

3-456

:lllExAs
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
N/C
20

RHPlN

CCRINRight 0.47 I1F
Negatlv~

23

Differential
Input
Signal

CRIN+
Right 0.47 11F
8
Positive
Differential
Input
SIHral
PCB EP
14
Input
Signal Cp B
0.4711F 2

-1

ROUT+

21

ROUT-

16

PVDD

18

VDD

19

BYPASS
SHUTDOWN

11

RIN

-::1

VDD

r~
-=-

Gain!
MUX
Control

CcLK
47nFT

Power
Management

Left CIIHP
Head- 0.4711F
phone
Input
Signal

-J

6

LHPIN

I..."--'WrlHJ:v:;;;;:=~.t----,G::.:N.::Dll

CILLINE
Left 0.4711F
Lina
Input
Signal

See Note A

1--'--'-"'-'<11---''''---,--- VDD
Depop
Circuitry

-J

CSR
1='0.1I1F
VOD

T

22

CSR
0.111F
CBYP

To 1=' 0.4711F
SystemControl

LOUT+

4

LOUT-

9

1,12,
13,24

1 kQ

COUTR
33OI1F

LIN
CLiN
0.47 11F

100kQ
NOTE A.

A 0.1 I1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 I1F or greater should be placed near the audio power amplifier.

Figure 31. Typical TPA0142 Application Circuit Using Differential Inputs

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3-457

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r------------

C
Input Signal

IN

I
I
I

Zf

ZI

----1f--.-..:.:.::..-I--'VItv-......
R

Figure 32, Resistor on Input for Cut-Off Frequency

The input resistance at each gain setting is given in Figure 28:
The -3 dB frequency can be calculated using the following formula:
f

1
-3 dB - 21t C(R II RI)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.

~TEXAS

3-458

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

TPA0142
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
input capacitor, C.
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input signal to the
form a
proper dc level for optimum operation. In this case, C, and the input impedance of the amplifier,
high-pass filter with the corner frequency determined in equation 2.

Z,.

fC(hiQhPaSS) =

(2)

2ltZ~NC,

The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where Z, is 710 kn and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
C -

1

I - 2ltZ,fc

(3)

In this example, C, is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 I1F. A further
consideration for this capacitor is the leakage path from the input source through the input network (C,) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
that the source dc 'evel. Note that it is important to confirm the capacitor polarity in the application.

power supply decoupling, Cs
The TPA0142 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 I1F placed as close as possible to the device VOO lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 I1F or greater placed near
the audio power amplifier is recommended.
mid rail bypass capacitor, CBYP
The mid rail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the mid rail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CBYP, values of 0.47 I1F to 1 I1F ceramic or tantalum low-ESR capacitors are recommended
for the best THO and noise performance.

~TEXAS

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3-459

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SL0S248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

(4)

fC(high)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 IlF is chosen and loads vary from 3 n,
4 n, 8 0, 32 n, 10 kn, and 47 kO. Table 2 summarizes the frequency response characteristics of each
configuration.

Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
RL

Cc

Lowest Frequency

30

330ILF

161 Hz

40

330ILF

120Hz
60Hz

ao

330ILF

320

330ILF

15 Hz

10,0000

330ILF

0.05 Hz

47,0000

330ILF

0.01 Hz

As Table 2 indicates, most of the bass response is attenuated into a 4-0 load, an 8-0 load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESR capaCitors are recommended throughout this applications section. A real (as opposed to ideal)
capaCitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capaCitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

~TEXAS

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TPA0142
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

- -_ •• - ..... """ ••••• ~"ft •• A"'I""I"".
A ...... L.I\"AIIUI.. lI .. rvn'.'''''"vn

bridged-tied load versus single-ended mode
Figure 33 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0142 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 5).
VO(PP)

V(nns) =

(5)

212
2

Power

V(nns)

F\
VDD

*
J'!

V'
RL
VDD

vO(PP)

2x VO(PP)

-=Figure 33. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-0 speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 34. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 ~F to 1000 ~F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
(6)

:ilTEXAS

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3--461

TPA0142
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B -JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
For example, a 68-~F capacitor with an 8-0 speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD

~dB~----~~====

Figure 34. Single-Ended Configuration and Frequency Response

Increasing power to the load does carry a penalty of increased intemal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Intemal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

single-ended operation
In SE mode (see Figure 33 and Figure 34), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 1 VN.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the intemal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The intemal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the intemal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMSand average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 35).

~TEXAS

3-462

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TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATiON
100

,/

---fVWV"ffll.-

V(LRMS)

IOO(avg)

Figure 35. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
P

Efficiency of a BTL amplifier = ~
SUP

(7)

Where:
VLrms2

PL

= ~'

andVLRMS

=

Vp

V 2

.[2' therefore, PL

and

looavg

=~

f"

= 2~L
V

0 R: sin(t) dt

=~

V

1t

x R: [cos(t)] 0

2V

= 1t ~

Therefore,
_ 2 Voo Vp
Psup 1t RL
substituting PL and Psup into equation 7,

v/

Efficiency of a BTL amplifier
Where:

~

1tVp
2Voo Vp = 4 Voo
1t RL

PL =Power delivered to load
Psup = Power drawn from power supply
VLRMS =RMS voltage on BTL load
RL = Load resistance
Vp =Peak voltage on BTL load
looavg =Average current drawn from
the power supply
VOO =Power supply voltage
l1BTL =Efficiency of a BTL amplifier

Therefore,

(8)

"BTL

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3-463

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B-JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 3. Efficiency Vs Output Power In 5-Va-0 BTL Systems
Output Power
(W)

Efficiency
(%)

Peak Voltage

0.25

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.4rt

0.53

(V)

Intemal Dissipation
(W)

t High peak voltages cause the THO to increase.
A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature the internal
disSipated power at the average output power level must be used. From the TPA0142 data sheet, one can see
that when the TPA0142 is operating from a 5-V supply into a 3-0 speaker that 4 W peaks are available.
Converting watts to dB:
P dB

=

P

10Log-Yt
Pref

=

10Log 4 W

1W

=

6 dB

(9)

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6 dB -15 dB = -9 dB (15 dB crest factor)
6 dB -12 dB = -6 dB (12 dB crest factor)
6 dB - 9 dB = -3 dB (9 dB crest factor)
6 dB - 6 dB = 0 dB (6 dB crest factor)
6 dB - 3 dB = 3 dB (3 dB crest factor)
Converting dB back into watts:
P

w=

10PdB/10 x P ref

= 63 mW (18 dB crest factor)
= 125 mW (15 dB crest factor)
= 250 mW (9 dB crest factor)

= 500 mW (6 dB crest factor)
= 1000 mW (3, dB crest factor)
= 2000 mW (15 dB crest factor)

-!111EXAS
INSTRUMENTS
3-464

POST OFACE BOX 655303 • DALLAS. TEXAS 75265

(10)

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-0 system, the intemal dissipation in the TPA0142 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0142 Power Rating, 5-V, 3-n, Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE
-3°C

POWER DISSIPATION

4

2W(3dB)

1.7

4

1000 mW (6 dB)

1.6

6°C

4

500mW(9dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

78°C

4

63 mW (18 dB)

0.6

96°C

Table 5. TPA0142 Power Rating, 5-V, 8-n, Stereo
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

1250 mW (3 dB crest factor)

0.55

100°C

1000 mW (4 dB crest factor)

0.62

94°C

2.5W

500 mW (7 dB crest factor)

0.59

97°C

2.5W

250 mW (10 dB crest factor)

0.53

102°C

PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

2.5W
2.5W

POWER DISSIPATION

The maximum dissipated power, POmax, is reached at a much lower output power level for an 8 0 load than for
a 3 0 load. As a result, this simple formula for calculating POmax may be used for an 8 0 application:

2Vfm
POmax =

(11 )

3t 2R

L

However, in the case of a 3 0 load, the POmax occurs at a pOint well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 0 load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to SJA:

e

JA

=

1
= _1_
Derating Factor
0.022

= 450CjW

(12)

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3-465

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given eJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0142 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
TA Max

=
=

T J Max - ElJA Po
150 - 45(0.6 x 2)

(13)

=

96°C (15 dB crest factor)

NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.

Tables 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0142 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
speakers dramatically increases the thermal performance by increasing amplifier
significantly. Also, using
efficiency.

a-n

SE/BTL operation
The ability of the TPA0142 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0142, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When
SE/BTL is held low, the amplifier is on and the TPA0142 is in the BTL mode. When SE/BTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0142 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SE/BTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 36.
20

RHPIN

23

RLINEIN

8

RIN

ROUT+

21

voo
ROUT-

16

100kn

sEim

15 100kn

~

n

.----~
Figure 36. TPA0142 Resistor Divider Network Circuit

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3-466

POST OFFICE BOX 655303 • DALLAS. TEXAS 7526Q

TPA0142
2-W STEREO AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL
SLOS248B - JUNE 1999 - REVISED MARCH 2000

• - ...............,...,.......
•• &"'1'"1"".
AI""I""'-""A I lUI' ",r",n.n,'" ."' ...
~"I"I

Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 100-knt1-kn divider pulls the SElBTL input low. When a plug is inserted, the 1-kn
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the OUT-amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.

PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is normally activated activated automatically, but may be
selected manually by pulling PCB ENABLE high. When the PC BEEP input is active, both of the LlNEIN and
HPIN inputs are deselected and both the left and right channels are driven in BTL mode with the signal from
PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is independent ofthe volume
setting. When the PC BEEP input is deselected, the amplifier will return to the previous operating mode and
volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC BEEP will take the device out
of shutdown and output the PC BEEP Signal, then return the amplifier to shutdown mode.
When PCB ENABLE is held low, the amplifier will automatically switch to PC BEEP mode after detecting a valid
signal at the PC BEEP input. The preferred input signal is a square wave or pulse train with an amplitude of 1
Vpp or greater. To be a accurately detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall
times of less than 0.1 J1S and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier
will return to its previous operating mode and volume setting.
When PCB ENABLE is held high, PC BEEP is selected and the LlNEIN and HPIN inputs are deactivated
regardless of the input signal. PCB ENABLE has an internal 100 kn pulldown resistor and will trip at
approximately Vool2.
If it is desired to ac couple the PC BEEP input, the value of the coupling capaCitor should be chosen to satisfy
the following equation:

C

>
1
PCB - 2lt fpCB (100 kQ)

(14)

The PC BEEP input can also be dc coupled to avoid using this coupling capacitor. The pin normally sits at mid rail
when no signal is present.

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3-467

TPA0142
2·W STEREO AUDIO POWER AMPLIFIER
WITH· DC VOLUME CONTROL
Sl0S248B -JUNE 1999 - REVISED MARCH 2000

·APPLICATION INFORMATION
Input MUX operation
Right
Headphone
Input
Signel

CIRHP
O.47 11F

--j

R
CIRLINE
OA711F

RlghtLlne
Input
Signal

23

MUX

RLiNEIN

~
~7

8

ROUT+

21

ROUT-

16

RIN

I
Figure 37. TPA0142 Example Input MUX Circuit

Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SElBTL operation section for a description of the headphone jack control circuit.

shutdown modes
The TPA0142 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, 100 = 150 ItA. SHUTDOWN. should never be left
unconnected because amplifier operation would be unpredictable.
Table 6. Shutdown and Mute Mode Functions
AMPLIFIER STATE

INPUTSt
SElBTL

SHUTDOWN

INPUT

Low

High

Line

BTL

X

Low

X

Mute

High

High

HP

SE

t Inputs should never be left unconnected.
X =do not care

3-468

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TEXAS
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OUTPUT

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
• Compatible With PC 99 Desktop Line-out
Into 10-k.Q Load
• Compatible With PC 99 Portable Into &-n
Load
• Internal Gain Control, Which Eliminates
External Galn-SeHlng Resistors
• Digital Volume Control From +20 dB to
-40 dB
• 2-W/Ch Output Power Into 3-n Load
• PC-Beep Input
• Depop Circuitry
• Stereo Input MUX
• Fully Differential Input
• Low Supply Current and Shutdown Current
• Surface-Mount Power Packaging
24-Pin TSSOP PowerPADlM

_....-....

...... ,. r"'''''''''''.... ~
-.,&_~

(TOP VIEW)

GND
UP
DOWN
lOUT+
lLiNEIN
lHPIN
PVOO
RIN
lOUTLIN
BYPASS
GND

10
2
3
4
5

6
7
8
9
10
11
12

24
23
22

21
20
19
18
17
16
15
14
13

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
VOO
PVOO
ClK
ROUTSElBTl
PC-BEEP
GND

description
The TPA0152 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-n loads. This device minimizes the number of
external components needed, which simplifies the design and frees up board space for other features. When
driving 1 W into 8-n speakers, the TPA0152 has less than 0.3% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause. noise in
the speakers.
The overall gain of the amplifier is controlled digitally by the UP and DOWN terminals. At power up, the gain
is set at the lowest level, -85 dB. It can then be adjusted to any of 31 discrete steps by pulling the voltage down
at the desired pin to logic low. The gain is adjusted in the initial stage of the amplifier as opposed to the power
output stage. As a result, the THD changes very little over all volume levels.
An intemal input MUX allows two sets of stereo inputs to the amplifier. In notebook applications, where intemal
speakers are driven as BTL and the line outputs (often headphone drive) are required to be SE, the TPA0152
automatically switches into SE mode when the SElBTL input is activated. This effectively reduces the gain by
a dB.
The TPA0152 consumes only 10 mA of supply current during normal operation. A miserly shutdown mode is
included that reduces the supply current to less than 150 J.IA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are truly realized in multilayer PCB
applications. This allows the TPA0152 to operate at full power into 8-n loads at ambient temperatures of 85°C.

~

~

aV~lIability,

Please be aware that an important notice conceming
standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of ·thls data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

Copyright © 2000, Texas Instruments Incorporated

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-469

TPA0152
2;.W STEREO AUDIO .POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

functional block diagram
RHPIN

~

RLINEIN - - - - - 1

M~X

...

~,.-..,....

>-....- - - - - - -

ROUT+

~~-I-------

ROUT-

UP-------.
DOWN

RIN

- - - - - - - - t -....
-------+--+---1--.

PC-BEEP --1L._e':_';_P---,

Power
Management
SElBTL

LHPIN

' - - - - - - - - GND

[;gM~X

LLINEIN - - - - i

LIN

...

~--

>-....- t - - - - - - -

LOUT+

>--.-------

LOUT-

-----------l-~

~TEXAS

INSTRUMENTS
3-470

PVDD
VDD
BYPASS
SHUTDOWN

POST OFFICE BOX 655303 • DAlLAS. TEXAS 75265

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

AVAiiAtii..i: OFriQNS
PACKAGED DEVICE

t

TA

TSSOpt
(PWP)

-40°C to 85°C

TPA0152PWP

The PWP package is available taped and reeled. To order a taped and reeled part,
add the suffix R to the part number (e.g., TPA0152PWPR).

Terminal Functions
TERMINAL
NAME

NO.

110

DESCRIPTION

BYPASS

11

ClK

17

I

If a 47-nF capacitor is attached, the TPA0152 generates an internal clock. An ex1ernal clock can override the
intemal clock input to this terminal.

DOWN

3

I

A momentary pulse on this terminal decreases the volume level by 2 dB. Holding the terminal low for a period
of time will step the amplifier through the volume levels at a rate determined by the capacitor on the ClK
terminal.

GND

Tap to voltage divider for internal mid-supply bias generator

1,12
13,24

Ground connection for circuitry. Connected to thermal pad

lHPIN

6

I

left-channel headphone input, selected when SElBTl is held high

LIN

10

I

Common left input for fully differential input. AC ground for single-ended inputs

lLiNEIN

5

I

left-channel line negative input, selected when SElBTl is held low

lOUT+

4

0

left-channel positive output in BTL mode and positive in SE mode

lOUT-

9

0

left-channel negative output in BTL mode and high impedance in SE mode

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > 1-V (peak-to-peak) square wave is input to
PC-BEEP or PCB ENABLE is high.
Power supply for output stage

PC-BEEP
PVDD

7, 18

I

RHPIN

20

I

Right channel headphone input, selected when SEIBTl is held high

RIN

8

I

Common right input for fully differential input. AC ground for single-ended inputs

RLiNEIN

23

I

Right-channel line input, selected when SElBTl is held low

ROUT+

21

0

Right-channel positive output in BTL mode and positive in SE mode

ROUT-

16

0

Right-channel negative output in BTL mode and high impedance in SE mode

SElBTl

15

I

Input MUX control input. When this terminal is held high, the lHPIN or RHPIN and SE output is selected. When
this terminal is held low, the lLiNEIN or RLiNEIN and BTL output are selected.

SHUTDOWN

22

I

When held low, this terminal places the entire device, except PC-BEEP detect circuitry, in shutdown mode.

UP

2

I

A momentary pulse on this terminal increases the volume level by 2 dB. Holding the terminal low for a period
of time will step the amplifier through the volume levels at a rate determined by the capacitor on the ClK
terminal.

VDD

19

I

Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-471

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL

SL0S246B- JUNE 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, VOD ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ....................• intemally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .............. • . . . . . . . . . . . . . . . . . . . . . . . . . .. -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/1Efinch) from case for 10 seconds ............................... 260°C
t Stresses beyond those listed under "absolute maximum ratings· may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions' is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
PWP

DERATING FACTOR

2.7 WI:

21.8mWrC

1.7W

1.4W

:j: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

(literature number SLMAOO2), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the befora mentioned document.

recommended operating conditions
Supply voltage, VOO
High-level input voltage, VIH
Low-level Input voltage, VIL

MAX

4.5

5.5

SElBTL

4

SHUTDOWN

2

SElBTL
SHUTDOWN

Operating free-air temperature, TA

INSTRUMENTS
POST oFFICE BOX 655303 • DALLAS, TEXAS 75265

UNIT
V
V

3
0.8
-40

~1ExAs

3-472

MIN

85

V
·C

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, V DD
noted)

=5 V, i A = ~5nc (Uii:6SS ..t~Q;'''·''::::C

TEST CONDITIONS

PARAMETER
IVool

Output offset voltage (measured differentially)

V,=O, AV=2

PSRR

Power supply rejection ratio

VOO=4.9Vt05.1 V

IIIHI

High-level input current

VOO=5.5V,
V,=VOO

IIILI

Low-level input current

VOO =5.5 V,
V,=OV

100

Supply current

IOO(SO)

Supply current, shutdown mode

MIN

TYP

MAX

UNIT

25

mV

67

BTL mode

10

SEmode

5
150

dB
900

nA

900

nA
mA
~A

300

operating characteristics, VDD =5 V, TA = 25°C, RL = 4 n, Gain = 2 VN, BTL mode (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

MIN

TYP

Po

Output power

THO = 1%,

f = 1 kHz

THO+N

Total harmonic distortion plus noise

PO=1 W,

f=20Hzt015kHz

BaM

Maximum output power bandwidth

THO = 5%

Supply ripple rejection ratio

f= 1 kHz,
CB =0.47 ~F

BTL mode

65

SEmode

60

CB =0.47 ~F,
f= 20 Hz to 20 kHz

BTL mode

17

Noise output voltage

SEmode

44

Vn

MAX

UNIT
W

2
0.3%

kHz

>15

dB

~VRMS

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power
THO+N

Total harmonic distortion plus noise

vs Gain
vs Frequency

Vn

SNR

2
3,5,7,9,11,
14

Output noise voltage

vs Frequency

13

Supply ripple rejection ratio

vs Frequency

14,15

Crosstalk

vs Frequency

16,17,18

Shutdown attenuation

vs Frequency

19

Signal-te-noise ratio

vs Frequency

20

Closed loop respone
Po

1,4,6,8,10,
12

21,22

Output power

Po

Power dissipation

ZI

Input impedance

vs Load resistance

23,24

vs Output power

25,26

vs Ambient temperature

22

vsGain

28

~TEXAS

INSTRUMENTS
POST OFFICE

eox 655303 •

DALLAS, TEXAS 75265

3-473

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S246B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE
va
GAIN

vs
OUTPUT POWER
10%

1%

I

1
I

+

i

1%

J

~
r-

I

J I

RL=4U

RL=3U

=
-

J

:!

0.1%

~

~

~

"'-

.!:!

S

0.1%

~

I

WforAV~B

Vo = 1 VRMS for Ays4 dB
r- RL=8U
BTL

I
I

I II

1

r::

+

I II

RL=8U

r- PO=1

I

I

--

I

z

0

~

-

AV = +20 to OdB
f=1kHz
Bn

:z:

I-

-

0.01%
0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 2.75

0.01%
-40

3

-30

Po - Output Power - W

Figure 1

'"

-20
-10
o
AV - Voltage Gain - dB

10

20

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

RL=3U
AV = +20 to 0 dB
Bn

Iz

+

c

~

J
~
!..

PO=1W

III

0

PO=0.5W
~

',",

r""

~1Ii

:z:

~

1%

r-I'oo
0.1% ~

I

'--

P? =1 \7~1~ 110.01 %
20

1k
f - Frequency - Hz

~I

./
RL=3U
AV = +20 to 0 dB
BTL

j:

0.01%
0.01

Figure 3

0.1
Po - Output Po_r - W

Figure 4

~I
'I TEXAS
3-474

"

f=20Hz

z

10k 20k

J

f=1 kHz

0

1111111

100

f=20kHz

NSTRUMENTS

POST OFFICE BOX 665303 • DALlAS, TEXAS 75265

10

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S2468 - JUNE 1999 - REVISED MARCH 2000

TYPICAL cHARACTERiSTiCS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

10%

.~

10%

+
c

+

c

0

'f

I

~

1%

I

.S!
c
Po= 0.25 W

:!

Iz

./

0.1%

:z:

Iz

PO=1.5W

I

+

j!:
0.01%
20

1=1 kHz
100..

0.1%

I

+

I"'"

Q

11= 20 kHz

-.. ~ ...

~

~--

.......... ....

1%

i

.-.;

0

I!!

RL=40
Ay = +20 to OdB
BTL

Iz

RL=40
Ay = +20 to 0 dB
BTL

~

1=20 Hz

Q

j!:

iii Ilrrl

100
1k
f - Frequency - Hz

1111

0.01%
0.01

10k 20k

Figure 5

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

~

~

I

~

c-

+

I..

10%
RL=80
Ay = +20 to OdB
BTL

Iz

-c:: RL=80

+

~

1%

J o.1% ~
~
z

~
0.01""
20

~

II

I
PO=0.5W

EIJ~lkHz I

~0
I!!

r-

:!
J

0.1%

~

1=1 kHz

t-..

I

--

Z

+

Q

j!:

PO=1W

100

I

1%

I
Po = 0.25 W

'"

,... Ay=+20toOdB
- BTL

c

J

10

0.1
Po - Output Power - W

1k
f - Frequency - Hz

10k 20k

0.01%
0.01

f=20Hz

1 IIIIII

Figure 7

0.1
Po - Output Power - W

10

FigureS

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-475

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10

10%
RL=32Q
AV=+14toOdB
SE

I
+

I

.;
z0

+

c

%

--

0.1

IS

~

~.

z
c!i
j!:

f=20kHz

1-0

0.1%

I

~f=1kHz

Z

PO=50mW
0.001I""
20

-

{!!.

0.01 %r'--

I

=

0

IS

Po=25mW

r--.

{!.

1%

i5
.!:!
c

~

.2

g

~
~

~

PO=75mW

r-.....

%

I-

11 IIII

IIIIIII
100

+

Q

1k
f - Frequency - Hz

0.01%
0.01

10k 20k

f=20Hz

0.1
Po - Output Power - W

Figure 9

Figure 10

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

FREQUENCY

10%

10%

~ RL=10kQ
~ AV=+14toOdB

.1
~

I

I- SE

+
c

+

1%

~

I

s
is

J

1%

-\ K

Q

~

.~ 0.1%

IS

VO=1 vRMS

t-...

~

If

u

0.1%

I
z

RL=32Q
AV = +14 to 0 dB
SE

I

0.001%
20

.

z
c!i
j!:
100

1k
f - Frequency - Hz

10k 20k

.~

PO=1 kHz

~. RL~10kfl
t- AV=+14toOdB

0.001%

r
o

PO=20Hz

SE

1 I

0.2 0.4 0.6

0.8

1 1.2 1.4 1.6 1.8

Vo - Output Voltage - VRMS

Figure 11

Figure 12

~TEXAS
3-476

~

0.010/0

I

oj!:

I

~

I-

{!.

0.01%

I
PO=20kHz

INSTRUMENTS

POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

2

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS
OUTPUT NOISE VOLTAGE

SUPPLY RIPPLE REJECTION RATIO

vs

vs

FREQUENCY
160
VI

140

~

120

:e
II:
I

GI
aJ

!

.~

~

0
I

0

I I III

BW = 22 Hz to 22 kHz
RL=40

III
'a

RL=SO
CB=0.47 IlF
BTL

-20

I

I

V

100

~

z
'S

I

-~~~~I~I~

FREQUENCY

II:

tlGI

SO

'Ii'

t

V"

Q.

ii:

Ay=+6dB

40

,/

>c

o

o

100

I'

"""

8:
:::I

VI

1k
f - Frequency - Hz

I\,

-80

~

if'"

Ay=+6dB

~

~

,

20

-60

II:

j...ooj..oo

60

-40

c
0

j..--~

AV=+20dB

i

111111
Ay=+20dB

-100

-120

10k 20k

20

100

1k
f - Frequency - Hz

Figure 13

Figure 14

SUPPLY RIPPLE REJECTION RATIO

CROSSTALK

vs

vs

FREQUENCY
0

III
'a

I

I

CB =0.471lf
-20 r- SE

I

-50

!'or'"

0

ic

-40

i

-60

-60

............

.2

i"-r--

.!!
Q.

.9- -80
II:

~

\

III
'a

...I

..e

~~ ~111 VJ

-70

!

-80

0

-90

I

RL=SO
Ay = +20 dB
Bn

/'
LEFT TO RIGHT

/v

t:...

"

V

v~

RIGHT TO LEFT

Ay=+14dB

~

g.

AV=+6dB

I~ P

II:

VI

FREQUENCY
-40

I~LI~ ~~IO

10k 20k

-100
-100

-110

-120
20

100

1k
f - Frequency - Hz

10k 20k

-120
20

100

1k

10k 20k

f - Frequency - Hz

Figure 15

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-477

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S2468 - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CROSSTALK
vs
FREQUENCY

CROSSTALK
vs
FREQUENCY
0

PO=1W
RL=80
AV=+6OdB
BTL

~

I:

-20

11llli

-70

/

LEFT TO RIGHT
II I ,

I

r-

VRMS

VO=l
RL=10kn
Av=+6dB
SE

~I~~rtj;f

...... 1..0-""

./

-40

ID

"I

I

-60
LEFT TO RIGHT

-60

~

-100

RIGHT TO LEFT
-100

-110
-120
20

100

lk
f - Frequency - Hz

-120
20

10k 20k

Figure 18

SHUTDOWN ATTENUATION
vs
FREQUENCY

SIGNAL·TO·NOISE RATIO
vs
FREQUENCY

0

120
VI=lVRMS

ilI'

-20
ID

i

ID

~

RL=10kn,SE

-40

"I

ta:
J

-60

~

RL=32o,SE

~

!
Q

-60

~

(-.

-100

105
100

f""

~f""

95

z
til

lL
100

lk
f - Frequency - Hz

10k 20k

t--

90
j

80

o

100

~y, ~ +6 ,dB,

IIIIII

lk

f - Frequency - Hz

Figure 19

Figure 20

~1ExAs

INSTRUMENTS
3-478

(-.,

85

111111111

AV= +20 dB

r--

r-- l-

I

a:

RL=8O, BTL
-120
20

110

US

I'r-

m

PO=lW
RL=80
BTL

115

~

j
I
;

10k 20k

f - Frequency - Hz

Figure 17

"CI

lk

100

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

r10k 20k

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERiSTiCS

CLOSED LOOP RESPONSE

30

1~~I~I~nl

IIII

II

25 ,.- AV = +20 dB
BTL

~~l~1

20
15
III
'1:1

~

~

10

IIII

~

"

1\

~~~~~

I-

!\

r\

5

~

o
--5

-10
10

1k

100

10k

-180°

100k

1M

f - Frequency - Hz

Figure 21

CLOSED LOOP RESPONSE

30

~ "'
RL=8n
AV=+6dB
BTt.

25
20
15
III
'1:1

~

~

10

'I"-

Phase
;1111

1111

5

t\

II

\~

GaIn

o

1\

--5

-10
10

100

1k

10k

100k

-180°
1M

f - Frequency - Hz

Figure 22

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-479

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL

SL0S246B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER
va
LOAD RESISTANCE

OUTPUT POWER
vs
LOAD RESISTANCE

3.5
3

i:

2.5

\

I

2

\

I

!is

t
0

1500

Ay=+20toOdB
BTL

1.5

Ay = +14 to 0 dB
1250 lJ

1

~
I

I

~ 10%THD+N

D.

i

\~

I

0

~

rP

~
0.5

J>

f::: ~

~

1000

750

~
\

500

250

1%THD+N

o

IIIII

o

SE

8

16
24
32
40
48
RL - Load Resistance - 0

56

o

64

~

10%THD+N

1%TH~
1
I

o

~

~
~
40
48
RL - Load Resistance - 0

8

POWER DISSIPATION
va
OUTPUT POWER

,v

1.6

i:

1.4

/
//
II

I

c:
0

I.

1.2

1
is

I V.,
0.8

I

,e

0.6
0.4

80

V

40

....

0.4

-

--

0.35

i:
I

0.3

i

0.25

c:

:ICI

0.2

r--.. ~o

I
Q
D.

~

1

1.5

2

1

............

Po - Output Power -

w

o
o

80

'"I'

0.1

320
0.05 ~
2.5

""

-'-

I rl'L -

f=1 kHz
BTL
Each Channel

0.5

......

/

0.15

0.2

o
o

POWER DISSIPATION
vs
OUTPUT POWER

30

/'
.".,-

64

Figure 24

Figure 23

1.8

56

f= 1 kHz

BTL
Each Channel

~

~

u

~

~

~

Po - Output Power - W

Figure 25

Figure 26

:lllExAs

INSTRUMENTS

POST OFFICE BOX 856303 • DALlAS, TEXAS 75285

u

U

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S246B -JUNE 1999 - REVISED MARCH 2000
-.-.~

•. _........

A"~"".I!'''.'''~

• 't"'''''L " " " " " " I ((;;,"I~ I I ....~

POWER DISSIPAnON
vs
AMBIENT TEMPERATURE
7

\

~
I

5

i

4

c

~

I
I

Q
Q.

=45.9°C/W
=45.2°C/W
=31.2°CIW
=18.6°C/W

-

1\

""~1\

jJA3

1

3

.....""

9JA1,2
2

o

9JA1
9JA2
9JA3
9JA4

\

9JA4
6

"- f' \

"1\
~~
........

~

-40 -20

0

20

40

80

'"

80 100 120 140 160

TA - Ambient Temperature - °C

Figure 27
INPUT IMPEDANCE
va
GAIN
90

80

.ilI

70

3c

80

t

.§
!i
a.
.5

" '\,
1\

50

\

40

I

N

30

\

20
10
-40

\

-30

-20
-10
AV-Galn-dB

o

10

"

20

Figure 28

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3-481

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
VDD
UP

100kn

O.471lF

YGND

~ 100kn

I - = - 2 UP
LOUT+i

~~

3 DOWN
0.471lF

LLINEO~-

0.471lF
II

LHPO~-

4 LOUT+

I~ LLiNEIN

0.47 1l F
1

6 LHPIN

,--18

I

9
LOUT-O~-

I

0.471lF

ri
0.47 IlF

10

*
-1

11
12

pVDD
RIN
LOUTLIN
BYPASS
GNO

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
VDD
PVDD
CLK
ROUTSElBTL
PC-BEEP
GND

I

24

~

0 R LINE

22

Shutdown

21
20

ROUT+

0.~7IlF

II

19

I

18

~~
16

t

0.11lF

0.11lF

:.J:- ~
T 101lF~
GND

-::-

f"'>

ROUT-

SElBTL

it---o PC-BEEP

14

IJ--

VOD

...L

15

13

1

0 RHP

0.47 1lF

Figure 29. Typical TPA0152 Application Circuit

selection of components
Figure 30 and Figure 31 are a schematic diagrams of typical notebook computer application circuits.

~TEXAS

INSTRUMENTS
3-482

POST OFFICE BOX 655303 • DALLAS, TEXAS 75255

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

Right CIRHP
Head- 0.47 ~F
phone
Input
SignaI
20
CIRLINE
Right 0.47~F
23
Line
Input
8
SignaI
CRIN
O.47~F T

--7

--7:

.J:-

PCB EEP IL14
Input-11
SignaI CPCB
0.47 ~F 17

n

RLiNEIN
RIN

3

100
kO

15

-.

'Uft?

R
MUX

-

UP
DOWN

ROUT+

.........

.~~

I

Gain!
MUX
SE!iffi: Control

If
:;;:::::: COUTR
330~F

ROUT-

100

PVDD
Depop
Circuitry

f-=-

Power
Management

-

Left CllHP
Head- 0.47 ~F
phone ~I
/I
Input
Sign al
CllLiNE
Left 0.47~F
Line
Input
Signal

~~

6
5

±
_

lLiNEIN

J
10

ClIN
0.47 ~F

lHPIN

LIN

-

18

VDD

19

BYPASS
SHUTDOWN

11
22

1

'~ft?

l'YD

VDD

111n

l--

VDD
CSR

'J:' 0.1 ~F

-1

VDD

-=-

~

CSR
0.1 ~F

1'J:'
CBYP
To 'J:' 0.47 ~F

System
Control
lOUT+

1--

?

..-

SaeNoteA

±

1 lin

I

GND

l
MUX

,A

16

100kO

lin

1--1

21

£J. }

ClK

47nFT

--2

-

PC-BEEP~
Bee

CCLI~-b

~-1
VOD

RHPIN

4

1,12,
13,24J:.

-

:::::=::: COUTR
~

330~F

-'

t

or>

II-....
lOUT-

9

100kO

~F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 ~F or greater should be placed near the audio power amplifier.

NOTE A. A 0.1

Figure 30. Typical TPA0152 Application Circuit Using Single-Ended Inputs and Input MUX

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2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL

SLOS246B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
N/C
Right
Negative
Differential
Input
20
SlgnaICCRINy 7 1lF 23
Right CRIN-.
Positive 0.47 IlF
Differently
8
Input
Signal
PC BEEP
14
Input
Signal Cpca
0.471lF 17

-1

RHPIN
R
MUX

RUNEIN

PC·BEEP

3
15

ROUT-

16

ClK

;-1

100

21

Ia':~

f]CClK
47nF'J

kO

ROUT+
RIN

DOWN
SE/aT

VDD
100

I-l
L:J

kO

Left -::- CIIHP
Head- 0.471lf
phona --11--+-,6"+""lH,,,P...,IN':'---I
Input
Signal
5
CllUNE r-J-!='=:!!!..-I
Left 0.471lF
Una
Input
Signal

PVDD 18 See Note A
1-...:.....:.==-1--'=----..,-- VDD
CSR

VDD
Power
Management BYPASS
SHUT·
DOWN

L__-JVI/Ir-+-.-~V:;:::::::I..J-----,G=N=Dll

-1

lOUT+

-:J' 0.11lF

19

-

11

VDD

'I'

22

CsR

0.11lF

Cayp

-:J' 0.47 jlf
SystemTo

1 kO

Control
112
4
13,24

COUTR

3301lf

UN
CUN
0.471lF
lOUT-

9

1ookO

NOTE A. A 0.1 IlF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency nOise signals. a larger
electrolytic capacitor of 10 IlF or greater should be placed near the audio power amplifier.

Figure 31. Typical TPA0152 Application Circuit Using Differential Inputs

~1ExAs

INSTRUMENTS
POST OFFICE BOX 656303 • DALlAS. TEXAS 75265

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000
... - - - - - - _ . _ . . . . . . . . """" ...... A"'rl"'"

At"t"LI~A.IIUI'I lI'\IIrUnIVU"IIVI'O

input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r------------

I
I
I

Rf

Input Signal ----1I---4I>--...;;:.;:.--I--Jl.J'V\r---I

R

The input resistance at each gain setting is given in the Figure 28.
The -3 dB frequency can be calculated using equation 1.

f
1
-3 dB - 21t C(R II RI)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.

Input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and the input impedance of the amplifier, ZIN, form a
high-pass filter with the corner frequency determined in equation 2.

fc(highpasS)

(2)

= 21ti, C I

The value of CI is important to consider as it directly affects the bass (lOW frequency) performance of the circuit.
Consider the example where ZI is 710 kQ and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.

C =_1_
I
21tZl fC

(3)

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TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL,
SL0S246B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Input capacitor, CI (continued)

In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 ~F. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
that the source dc level. Note that it is important to confirm the capacitor polarity in the application.
power supply decoupllng, Cs

The TPA0152 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 ~F placed as close as possible to the device Voo lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capaCitor of 10 ~F or greater placed near
the audio power amplifier is recommended.
mldrail bypass capacitor, CSyp

The mid rail bypass capacitor, CSyp, is the most critical capacitor and serves several important functions. Ouring
startup or recovery from Shutdown mode, CSyp determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THO+N.
Bypass capacitor, CSyp, values of 0.47 ~F to 1 ~F ceramic or tantalum low-ESR capacitors are recommended
for the best THO and noise performance.

output coupling capaCitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cd is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

fC(high)

(4)

fe

The main disadvantage, from a performance standpOint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. ConSider the example where a Cc of 330 ~F is chosen and loads vary from 3 n,
4 n, 8 n, 32 n, 10 kn, and 47 kn. Table 1 summarizes the frequency response characteristics of each
configuration.

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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S246B - JUNE 1999 - REVISED MARCH 2000

APPLiCATiON iNFORiviAiiOn
Table 1. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
RL

Cc

LOWEST FREQUENCY

30

33DIlF

161 Hz

40

3301lF

120Hz

ao

330llF
330llF

60Hz

320
10,0000

330llF

0.05 Hz

47,0000

330llF

0,01 Hz

15Hz

As Table 1 indicates, most of the bass response is attenuated into a 4-n load, an 8-n load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESA capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

bridged-tied load versus single-ended mode
Figure 34 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA0152 BTL amplifier
consists of two class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 5).

v

_

(rms) -

VO(PP)
2/2

(5)

2
V(rms)

-RL

Power -

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3-487

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS2468 - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Voo

J' :

J'!

RL
Voo

'V :

VO(PP)

2x vO(PP)

-VO(PP)

Figure 32. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-n speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concems.
Consider the single-supply SE configuration shown in Figure 33. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33J.1F to 1000 J.1F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
fe

=

(6)

1
2nRL C c

For example, a 68-J.1F capacitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
Voo

~dB~----~~====

fe

Figure 33. Single-Ended Configuration and Frequency Response

~TEXAS

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INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S246B - JUNE 1999 - REVISED MARCH 2000

AppliCATiON iNFORiviATiON
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor section.

single-ended operation
In SE mode (see Figure 32 and Figure 33), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 1 VN.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The internal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 34).
100

,/

--fVVVVVWl-

V(LRMS)

IOO(avg)

Figure 34. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

P

Efficiency of a BTL amplifier = ~
SUP

(7)

Where:
V L rms 2

Vp

P L = - - - andVLRMS
RL

and

Voo looavg and

.f2'

V 2

therefore, P L

= 2~L

f

1 It V P
looavg = it 0 RL sin(t) dt

=

1
VP
:rt
it x RL [cos(t)] 0

2V p
RL

=:rt

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-489

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Therefore,
_ 2 VOO Vp
PSUP n RL
substituting PL and Psup into equation 7,
V 2

Efficiency of a BTL amplifier

P
2RL

2Voo Vp
Where:

n RL

j2 PL RL

Vp
Therefore,

j2

PL RL
_ n
l1BTL 4 Voo

(8)

PL = Power devilered to load
PSUP = Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
Vp = Peak voltage on BTL load
looavg =Average current drawn from the power supply
VOO Power supply voltage
l1BTL = Efficiency of a BTL amplifier

=

Table 2 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo t-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 2. Efficiency vs Output Power In 5 V 8-0 BTL Systems
0

OUTPUT POWER

EFFICIENCY

PEAK VOLTAGE

(W)

(%)

(V)

(W)

0.25

31.4
44.4

2.00

0.55
0.62

0.50
1.00
1.25

62.8
70.2

INTERNAL DISSIPATION

2.83
4.00
4.47t

0.59
0.53

t High peak VOltages cause the THO to increase.

A final point to remember about class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

~TEXAS

3-490

INSTRUMENTS
POST OFFICE BOX 655300 • DAUAS. TEXAS 75265

TPA0152
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

AppliCATiON iNFORiviATiON
crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature the internal
dissipated power at the average output power level must be used. From the TPA0152 data sheet, one can see
that when the TPA0152 is operating from a 5-V supply into a 3-n speaker that 4 W peaks are available.
Converting Watts to dB:
P dB

=

10Log

(=:1) =

10L09(i~) = 6 dB

(9)

Subtracting the headroom restriction to obtain the average listening level without distortion yields:

6 dB -15 dB = -9 dB (15 dB crest factor)
6 dB - 12 dB =-6 dB (12 dB crest factor)
6 dB - 9 dB = -3 dB (9 dB crest factor)
6 dB - 6 dB = 0 dB (6 dB crest factor)
6 dB - 3 dB =3 dB (3 dB crest factor)
Converting dB back into watts:

Pw

=

10PdB/10 x Prel

=

63 mW (18 dB crest factor)

(10)

= 125 mW (15 dB crest factor)
= 250 mW (9 dB crest factor)

= 500 mW (6 dB crest factor)
= 1000 mW (3 dB crest factor)
= 2000 mW (15 dB crest factor)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-n system, the internal dissipation in the TPA0152 and
maximum ambient temperatures is shown in Table 3.
Table 3. TPA0152 Power Rating, 5-V, 3-0., Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE
-3°C

4

2W(3dB)

1.7

4

1000 mW (6 dB)

1.6

6°C

4

500mW(9dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

78°C

4

63 mW (18 dB)

0.6

96°C

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3-491

TPA0152
2..WSTEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S246B- JUNE 1999 - REVISEO MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations (continued)
Table 4. TPA0152 Power Rating, 5-V, 8-0., Stereo
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

1250 mW (3 dB crest laCIer)

0.55

100°C

1000 mW (4 dB crest laCIer)

0.62

94°C

2.5W

500 mW (7 dB crest lactor)

0.59

97"C

2.5W

250 mW (10 dB crest laCIer)

0.53

102°C

PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

2.5W
2.5W

POWER DISSIPATION

an

The maximum dissipated power, POmax, is reached at a much lower output power level for an
load than for
a 3 n load. As a result, this simple formula for calculating POmax may be used for an a n application:

2Vr>D
P omax = :n;2R
L

(11)

However, in the case of a 3 n load, the POmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 n load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dispation rating table on page 4. Converting this to SJA:
El

JA

= Derating1 Factor = _1_
= 450C/W
0.022

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given SJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0122 is
150°C. The intemal dissipation figures are taken from the Pow&r Dissipation vs Output Power graphs.
T A Max = T J Max - ElJA Po

=

150 - 45(0.6 x 2)

(13)

=

96°C (15 dB crest factor)
NOTE:

Intemal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.
Tables 3 and 4 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0152 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 3 and 4 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using a-n speakers dramatically increases the thermal performance by increasing amplifier
efficiency.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, 1EXAS 75265

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S246B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
SE/BTL operation
The ability of the TPA0152 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0152, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When
SEIBTL is held low, the amplifier is on and the TPA0152 is in the BTL mode. When SEIBTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0152 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SElBTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 35.

20

RHPIN

23

RLiNEIN

8

R
MUX
ROUT+

21

ROUT-

16

RIN

Figure 35. TPA0152 Resistor Divider Network Circuit
Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-knl1-kO divider pulls the SElBTL input low. When a plug is inserted, the 1-kn
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the OUT-amplifier is
shutdown causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

3-493

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S246B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few extemal components. The input is normally activated activated automatically, but may be
selected manually by pulling PCB ENABLE high. When the PC BEEP input is active, both of the LlNEIN and
HPIN inputs are deselected and both the left and right channels are driven in BTL mode with the signal from
PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is independent of the volume
setting. When the PC BEEP input is deselected, the amplifier will retum to the previous operating mode and
volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC BEEP will take the device out
of shutdown and output the PC BEEP Signal, then retum the amplifier to shutdown mode.
When PCB ENABLE is held low, the amplifier will automatically switch to PC BEEP mode after detecting a valid
signal at the PC BEEP input. The preferred input signal is a square wave or pulse train with an amplitude of 1
Vpp or greater. To be a accurately detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall
times of less than 0.1 ~ and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier
will retum to its previous operating mode and volume setting.
When PCB ENABLE is held high, PC BEEP is selected and the LlNEIN and HPIN inputs are deactivated
regardless of the input Signal. PCB ENABLE has an intemal1 00 kn pulldown resistor and will trip at
approximately Vool2.
If it is desired to ac couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
the following equation:

c PCB ~ 231: fpCB 1(100

(14)

kg)

The PC BEEP input can also be dc coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.

~TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0152
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS246B - JUNE 1999 - REVISED MARCH 2000

APPLiCATiON iNFORiviATiOn
Input MUX operation
CtRHP
RIght
Headphone
Input
SIgnal

OA7 !1f

---1

R
CIRLINE
O.47 11F
RlghtLlne ~
Input
--;
Slgnel

23

8
CRIN
OA7!1f

MUX

RLINEIN

ROUT+

21

ROUT-

16

RIN

T
Figure 36. TPA0152 Example Input MUX Circuit

Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SE/BTL operation section for a description of the headphone jack control circuit.

shutdown modes
The TPA0152 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, Ipp = 150 J.LA. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.
Table 5. Shutdown and Mute Mode Functions
AMPLIFIER STATE

INPUTSt
SEfBTL

SHUTDOWN

INPUT

Low

High

Line

OUTPUT

BTL

X

Low

X

Mute

High

High

HP

SE

t Inputs should never be left unconnected.

=

X do not care

~lEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-495

3-496

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED

• Compatible With PC 99 Desktop Line-Out
Into 10-kO Load
• Compatible With PC 99 Portable Into 8-0
Load
• Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
• Digital Volume Control From 20 dB to
-40 dB
• 2-W/Ch Output Power Into 3-0 Load
• PC-Beep Input
• Depop Circuitry
• Stereo Input MUX
• Fully Differential Input
• Low Supply Current and Shutdown Current
• Surface-Mount Power Packaging
24-Pin TSSOP PowerPADTM

2000

..........
""'''''''.'._r"
r"'" .. r-r-""",,,,,,,"0;1'"
(TOP VIEW)

GND
UP
DOWN
lOUT+
lLiNEIN
lHPIN
PVDD
RIN
lOUTLIN
BYPASS
GND

7

24
23
22
21
20
19
18

8
9
10
11
12

16
15
14
13

10
2
3
4
5
6

17

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
VDD
PVDD
ClK
ROUTSElBTl
PC-BEEP
GND

description
The TPA0162 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-0 loads. This device minimizes the number of
external components needed, which simplifies the design and frees up board space for other features. When
driving 1 W into 8-0 speakers, the TPA0162 has less than 0.22% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
The overall gain of the amplifier is controlled digitally by the UP and DOWN terminals. At power up, the gain
is set at the lowest level, -85 dB. It can then be adjusted to any of 31 discrete steps by pulling the voltage down
at the desired pin to logic low. The gain is adjusted in the initial stage of the amplifier as opposed to the power
output stage. As a result, the THO changes very little over all volume levels.
An internal input MUX allows two sets of stereo inputs to the amplifier. In notebook applications, where internal
speakers are driven as BTL and the line outputs (often headphone drive) ar~ required to be SE, the TPA0162
automatically switches into SE mode when the SElBTL input is activated. This effectively reduces the gain by
6 dB.
The TPA0162 consumes only 20 rnA of supply current during normal operation. A miserly shutdown mode is
included that reduces the supply current to less than 150 IJA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°CIW are truly realized in multilayer PCB
applications. This allows the TPA0162 to operate at full power into 8-0 loads at ambient temperatures of 85°C.

•

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments InCOrporated.

~~conr.:::l:=:'~~:::,e::=.:::

IIIndardwarranty. Production _sing does not necessarllf Include
testing 01 all paramol2ll.

~TEXAS

Copyrlght © 2000, Texas Instruments Incorporated

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-497

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

functional block diagram

~

RHPIN
RLiNEIN _ _ _

M~X

1..-..-........1

>--+-------

ROUT+

>-...-+-------

ROUT-

UP-------.
DOWN-------+--.
RIN -------+---t---+_~

PC-BEEP

--1. ::;~
Power
Management

SE/BTL

LHPIN

M~X

BYPASS
SHUTDOWN

' - - - - GND

g-

LLiNEIN _ _- j

PVDD
VDD

L..-_ _.....

>--+-1-------

LOUT+

>--+-------

LOUT-

LIN - - - - - ' - - - - - - - 1 - - - 4 1

~TEXAS

INSTRUMENTS
3-498

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S2498 - JUNE 1999 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICE
TA

TSSOP't
(PWP)

-40°C to 85°C

TPA0162PWP

t The PWP package IS available taped and reeled. To order a taped and reeled part,
add the suffix R to the part number (e.g., TPA0162PWPR).

Terminal Functions
TERMINAL
NAME
NO.

110

DESCRIPTION

BYPASS

11

ClK

17

I

If a 47-nF capacitor is attached, the TPA0162 generates an intemal clock. An extemal clock can override
the intemal clock input to this tenninal.

DOWN

3

I

A momentary pulse on this tenninal decreases the volume level by 2 dB. Holding the terminal low for a period
of time will step the amplifier through the volume levels at a rate detennined by the capac~or on the ClK
tenninal.

GND

Tap to voltage divider for intemal mid-supply bias generetor

1,12
13,24

Ground connection for circuitry. Connected to thennal pad

lHPIN

6

I

left-channel headphone input, selected when SElBTl is held high

LIN

10

I

Common left input for fully differential input. AC ground for single-ended inputs

lllNEIN

5

I

left-channelline negative input, selected when SElBTl is held low

lOUT+

4

0

left-channel positive output in BTL mode and positive in SE mode

lOUT-

9

0

left-channel negative output in BTL mode and high impedance in SE mode

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > 1-V (peak-te-peak) square wave is input to
PC-BEEP or PCB ENABLE is high.

7,18

I

Power supply for output stage

20

I

Right channel headphone input, selected when SElBTl is held high

RIN

8

I

Common right input for fully differential input. AC ground for single-ended inputs

PVDD
RHPIN
RLiNEIN

23

I

Right-channel line input, selected when SElBTl Is held low.

ROUT+

21

0

Right-channel positive output in BTL mode and positive in SE mode

ROUT-

16

0

Right-channel negative output in BTL mode and high impedance In SE mode

SElBTl

15

I

Input MUX control input. When this tenninal is held high, the lHPIN or RHPIN and SE output is selected.
When this tenninalls held low, the lLiNEIN or RLiNEIN and BTL output are selected.

SHUTDOWN

22

I

When held low, this tenninal places the entire device, except PC-BEEP detect circuitry, in shutdown mode.

UP

2

I

A momentary pulse on this tenninal increases the volume level by 2 dB. Holding the terminal low for a period
of time will step the amplifier through the volume levels at a rete detennlned by the capacitor on the ClK
terminal.

VDD

19

I

Analog VDD input supply. This tenninal needs to be isolated from PVDD to achieve highest performance.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 855303 • OAUAS, TEXAS 75265

3-499

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S249B - JUNE 1999 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ........................................................................ 6 V
Input VOltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
t Stresses beyond those listed under "absolute maximum ratings· may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those Indicated under "recommended operating conditions· is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

DISSIPATION RATING TABLE
DERATING FACTOR

PACKAGE
PWP

2.7W*

21.8mW/"C

1.7W

1.4W

:j: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VDD
High-level input voltage, VIH
Low-level input voltage, VIL

MIN

MAX

4.5

5.5

SElBTL

4

SHUTDOWN

2

SHUTDOWN

Operating free-air temperature, TA

0.8
-40

-!!1TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

V
V

3

SElBTL

UNIT

85

V
°C

TPA0162
2·W STEREO AUDIO. ,'IllER AMPLIFIER
WITH
DIGITAL
'I,,"1999.UME
CONTROL
__________________________
_ _siiiLiiio.S249iiiiiiB
..-.J_UNE
REVISED
MARCH 2000
electrical characteristics at specified free-air temperature, YDD
noted)
PARAMETER

=5 V, TA =2S"C (uniei5i5 other-wise

TEST CONDITIONS

MIN

TYP

h./=2

MAX

UNIT

IVool

Output offset voltage (measured differentially)

VI =0,

PSRR

Power supply rejection ratio

VOO =4.9Vt05.1 V

IIIHI

High-level input current

VOO=5.5V,
VI=VOO

900

nA

IIILI

Low-level input current

VOO=5.5V,
VI=OV

900

nA

100

Supply current

IOO(SO)

Supply current, shutdown mode

operating characteristics, VDD
noted)

mV

67

BTL mode

20

SEmode

10
150

dB

rnA

j.LA

300

=5 V, TA =25°C, RL =4 n, Gain =2 VIV, BTL mode (unless otherwise

PARAMETER

TEST CONDITIONS

Po

Output power

THO = 1%,

f=lkHz

THO+N

Total harmonic distortion plus noise

PO= 1 W,

f=20 Hz to 15kHz

BOM

Maximum output power bandwidth

THO=5%

Vn

25

MIN

TYP

MAX

UNIT
W

2
0.22%

kHz

>15

Supply ripple rejection ratio

f= 1 kHz,
CB=0.47 I1F

Noise output voltage

CB = 0.47 I1F,
f=20Hzt020kHz

BTL mode

65

SEmode

60

BTL mode

17

SEmode

44

dB

I1V RMS

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power
vsGain

1,4,6,8,10
2

THO+N

Total harmonic distortion plus noise

Vn

Output noise voltage

vs Bandwidth

13

Supply ripple rejection ratio

vs Frequency

14,15
16, 17, 18

vs Frequency
vs Output voltage

3,5,7,9,11
12

Crosstalk

vs Frequency

Shutdown attenuation

vs Frequency

19

SNR

Signal-to-nolse ratio

vs Bandwidth

20

Po

Output power

Closed loop respone

Po

Power dissipation

ZI

Input impedance

21,22
vs Load resistance

23,24

vs Output power

25,26

.vs Ambient temperature

27

vsGain

28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-501

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS2498 - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

VOLTAGE GAIN

10%

1%

Iz

~u

I

1

+

c

~0

/

.1

I

1%

t==
f--

'c0

RL=4Q!

J:

I
!J

RL=8Q

=
-

I I

I II

0.1%

~I

+

Ay = +20 to 4 dB
f = 1 kHz
BTL

~
0.01%
0.5 0.75

1

1.25 1.5 1.75 2

I

i

r-

t- RL=8Q

-

2.25 2.5 2.75

BTL

~

~

is

.Ii!

5 0.1%
E
!
S

'z7

Z

C

t- Po = 1 W for Ay>6dB

:= YO = 1 YRMS for A\F-4 dB

+

RL=3Q

i

S

L

............
r-...

~

~

~
3

0.01%
-40

-30

Po - Output. Power - W

Figure 1

-20
-10
o
Ay - Yoltage Gain - dB

10

20

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

RL=3Q
Ay = +20 to 0 dB
BTL

RL=3Q
Ay = +20 to +4 dB
BTL

.~

z

+

c

~
PO=0.5W ~
~

PO=lW

1%

is
.Ii!
c
0

r- t'-I'!!o.

~
I

PO=1.75W -

0.1%

k~ZI

1"V

~ -..

V

f=1

~ E:::

f:20Hz

J:

S

f=20kHz

IIIiII:

i

~~

".

~

t::::: t=

Z
+
C
J:

I-

0.01 %
20

1111111
100

1k
f - Frequency - Hz

10k 20k

0.01%
0.01

Figure 3

Figure 4

~TEXAS

3-502

0.1
Po - Output Power - W

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

10

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

TypiCAL CHARACTERiSTiCS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10"k

z

+
c

~0

.!z

Rl=4Q
Av = +20 to +4 dB
BTL

=

'0

+
c

~0

1%

~

..

~

~

E
-

I

:f"

RL=80
CB=0.47J1F
BTL

RL=40

8:::>

III

20~

IIIII

01-

o

100

1k
BW - Bandwidth - Hz

AV=+6dB

-100

11111

-120

10k 20k

20

100

Figure 13

1k
f - Frequency - Hz

Figure 14

SUPPLY RIPPLE REJECTION RATIO

CROSSTALK

vs

vs

FREQUENCY

o

FREQUENCY
-40

'RL'~ 32'0

'

-so

CB=0.47J1F

"oI

!

I..

"i"
-40

r-o."",

AViodB

-60

......

a:

i

RL=80
AV = +20 dB
BTL

-60

.........

-60

m

~

~I--

~

If

-60
-90

./

III I

-70

I

U

\
AV=+14dB

i

"

T

~~~'1 W

,

t- SE

m

10k 20k

~~

LEFT TO RIGHT ~

V

1-

I-'"i""

-

RIGHT TO LEFT

-100

::> -100

III

-110
-120
20

100

1k
f - Frequency - Hz

10k 20k

-120
20

100

1k

10k 20k

f - Frequency - Hz

Figure 15

Figure 16

:II
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-505

TPA0162
2·WSTEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CROSSTALK
vs
FREQUENCY

CROSSTALK
vs
FREQUENCY

-40
PO=1W
RL=80
AV=+6dB
BTL

-50
-60
ID

1e
u

IIllll

-70

'a
I

1

i

-60 J....I;.H+ttIt-+++ LEFT TO RIGHT

!

-80

...

"

-110
-120
20

I"'"

RIGHT TO LEFT

I

-70 H-+t+ttlt-+-t-++ttftt--+-+-t-t-ttHt---i

I

IJ

11111

-100

~

L~~TOIRIGHT

-60
-80 t-

VO=1 VRMS
RL=10kO
AV=+6dB
SE

-50

100

f...
1 1111
1~1t~~~~~I~lll~llll~+=I~~~~

I'
1"
RIGHT TO LEFT
-100 1-++++++++--+-"-'-TTTrn--,--I-H-ttttt---t
-110 H-t++tttt--t-t-+++tHt--+-t-t-ttt+tt-----i

1k

-120 L......L...L...LJ.JJ.II...-....L...J.....L...........~--'-..................~~
20
100
1k
10k 20k
f - Frequency - Hz

10k 20k

f - Frequency - Hz

Figure 17

Figure 18

SHUTDOWN ATIENUATION
vs
FREQUENCY

SIGNAL·TO-NOISE RATIO
vs
BANDWIDTH

0

120
VI=1 VRMS

-20
,

ID

'a
I

IS

IiII

~
I.c

ID

RL = 10 IUl, SE

~

-60

~

RL=32Q,SE

1"""

II)

~

I
0

110

,;0

105

z

/!.

100

!

II

95

II:

90

I

-40

-60
-100
-120
20

PO=1W
RL=80
BTL

115

!c

!

t--

-

~

II

Jj

f'... ro-.
1......

1"-

AV= +20 dB

t-- ro-.,...

...... t'--

....

AV=+6 dB

-""

r---....

85

Rriililili
100

r-....

1k

10k 20k

80

o

f - Frequency - Hz

1k
100
BW - Bandwidth - Hz

Figure 20

Figure 19

~lExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10k 20k

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE

30

111111

25 _

1SOO

IIII

RL=80
AV=+2OdB
BTL

~~:~I

20

ID

"I

~

15

'l
I

Phase

~

10

~I'I

5

o

-10
10

_180 0

100

10k

1k

100k

1M

f - Frequency - Hz

Figure 21

CLOSED LOOP RESPONSE

30

1SOO

"~III

RL=80
AV=+6dB
BTL

25

900

20
ID

15

~

10

"I

~

Phase
illll

I

11111

5

~

t\

,,~

Gain

o

-10
10

1\
_1800

100

1k

10k

100k

1M

f - Frequency - Hz

Figure 22

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75285

3-507

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999

REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

vs

vs

LOAD RESISTANCE

LOAD RESISTANCE

3.5

1500

AV= +20 to 0 dB
BTL

3

\

1250

2.5

==I

I

\

2

I

~\

'5

!

==E

1.5

I

10%THD+N

a.

...

'5
'5

\~

I

0

i\:

0

a.

~
"
IIIII

t-- r--..

0.5

o

I

~

AV=+14toOdB
SE

,

1000

750

~

500

~~

250

~

1%THD+N

S

o

10%THD+N

1%THD+N

16
24
32
40
48
RL - Load Resistance - n

56

o

64

~

o

S

I

16
24
32
40
4S
RL - Load Resistance - n

Figure 23

POWER DISSIPATION

vs

vs

OUTPUT POWER

OUTPUT POWER

-

1.S

==I

1.4

i
!
I

1.2

c

L~

a.
I

o.s
0.6

Q

a.

0.4

I

3n

lL'
~

//f'

0.4
0.35

sn

---

i

0.25

is

0.2

J

--

0.5

L
V

2

IL ~ r-..... '" sn
1/
"-l'

0.15
0.1

32n

0.05
2.5

r-..t

o
o

'=1kHz
BTL
Each Channel

I'

~

u

~

M

~

MUM

Po - Output Power - W

Figure 26

Figure 25

~TEXAS

INSTRUMENTS
3-508

f'...

/

Q

1.5
Po - Output Power - W

r-....... , n

10-'"'"

I

I

a.
'=1 kHz
BTL
Each Channel

0.2

o
o

0.3

0;

~

V

==I
c

4n

11

VL

64

Figure 24

POWER DISSIPATION

1.6

56

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
POWER DISSIPATION

vs
AMBIENT TEMPERATURE
7

I

6

1\

~
I

c

5

I

4

J
I

Q

"

jJA3,
3

9JA1,2
2

A.

o

-40 -20

1__ 1

\

0

!

I

9JA1 =45.9°CIW
9JA2 =45.2°CIW _
9JA3 =31.2°CIW
9JA4 = 18.6°CIW

\

9JA4

~

\

"""" ~

""

1\,

~~ \

"""" ~

,

0 20 40 60 80 100 120 140 160
TA - Ambient Temperature - °C

Figure 27
INPUT IMPEDANCE

vs
GAIN

90

80

~

70

I

CD

u

60

11Co

50

c
as

S

-- "'"

:;
Co

.5

"

\

40

\

I

N

30

\

20

10
-40

\

-30

-20
-10
AV-Gain-dB

o

10

"

20

Figure 28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-509

TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B-JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
VDD
UP

OA7 1lF

1-1------0 R LINE
GND

100 kf.l

-=-

UP

3
lOUT+i

4
0.47 1lF
lllNEO

0.47 1lF

II

lHPO

0.47 1lF

LOUT-O

0~
L

I

I~

DOWN

8
9
10

12

SHUTDOWN

LOUT+

ROUT+

lLiNEIN

RHPIN

6 lHPIN
7
PVDD

11

0.471lF

RLiNEIN

RIN

VDD
PVDD
ClK

LOUTliN

ROUTSEIBTL

BYPASS
GND

PC-BEEP
GND

22

Shutdown

21
20

ROUT+

0.471lF

I

o

RHP

19
18
17
16

47nF

h -=-

0.1 p.F

T

-=-

VDD

10p.F'T:........o
GND
ROUT-

15

SElBTL

14

~

13

PC-BEEP

0.471lF

-=-

-=-

Figure 29. Typical TPA0162 Application Circuit

selection of components
Figure 30 and Figure 31 are a schematic diagrams of typical notebook computer application circuits.

,

~TEXAS

3-510

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B- JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Right CIRHP
Head- 0.4711F
phone
Input
Signal
20 RHPIN
CIRLINE
R
Right OA711F
MUX . .----'V\/'~t--i..__'I/I.IIr---,
LIne
1----=2:;::3+-'R.::L:::IN.::E::.:IN~
Input
8
RIN
Signal

-j
-j

CRIN
0.4711F

T

ROUT+

21

ROUT-

16

I:.;~

PC BEEP
-::"
Input
14 PC-BEEP
Signal CPCB
0.47!1F ,.,..17,-+":'=:'>-_-,

---1

:-1n

CcLK

6

100
kn

47

nF~2
3
15

UP
DOWN
SEtB

Gain!
MUX
Control

l00kn

VDD

100
kn

PVDD 18 See Note A
t-..=....<.=-t--'=--,.-- Voo
CSR

Depop
ClrcuHry

VDD

19

Management BYPASS
SHUTDOWN

11

-:J' O.II1F
-

Power
CILHP
Head- 0.4711F
phone -11--+--,64-.!:!!LH~PI:!!Nl!..--I
Input
Signal
CILLINE .-"-1--"'==--1
Left 0.47 11F
Line -1
Input
Signal
Left

II-<.---'V\/'~t--i~:v::;;;::::::;-r.J--G::.:Nc.::Dll
LOUT+

VDO
CSR

-r O.II1F

22

CBYP
To

-:J' 0.4711F

System-

Control
1,12
4
13,24

1 kg

COUTR
33OI1F

LIN

LOUT-

9

l00kn
NOTE A. A 0.1 I1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals. a larger
electrolytic capacitor of 10 I1F or greater should be placed near the audio power amplifier.

Figure 30. Typical TPA0162 Application Circuit Using Single-Ended Inputs and Input MUX

~TEXAS

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TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL

SLOS249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
N/C
Right
Negative
DlfferenUal
20 RHPlN
Input
SlgnaICCRIN-'"""'+''''-''-=--I

I 0.~71-11_F13~~~4
'---j
RLiNEIN

R
MUX

Right CRIN+
Positive 0.47 11F
Differential ~1-=-8-t-=R=IN~_ _ _..,
Input
I
Signal

J

I

--1

ROUT+

21

ROUT-

16

~

PC BEEP
14 PC-BEEP BeePCInput
Signel CPCB
0.47 j1f "'17......",C""LK"-_--,
CCLK
~~6 47nF'.J;
:),
-2 UP
Gain!
3 DOWN
100
MUX
kn
15 SElBT Control
VDD

II

100
kn

100kn
PVDD
Depop
Circuitry

l-1f

Power
Management

Left -=- CIIHP
Head- 0.4711F

phone
Input
Signal

-11--+-'6. . . . .=LHPI-"N"---t

VDD

19

BYPASS
SHUT·
DOWN

11

-1

22

VDD

-:f 0.1CSR

j1f

VDD

T

P

CSR
0.111F

-=-

-::r

LOUT+

CBYP
To
0.4711F
System
Control
112
13,24
4

LOUT-

9

GND

c.:.

CILLINE r"--1--"'=:!"'---1
Left 0.47 11F
Line
Input
Signal

18 See Note A

-=-

1kn

-=-

COUTR
330 j1f

LIN
CLiN

0.47 11F

-=-

100kn
NOTE A. A 0.1 I1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower·frequency noise signals, a larger
electrolytic capacitor of 10 I1F or greater should be placed near the audio power amplifier.

Figure 31. Typical TPA0162 Application Circuit Using Differential Inputs

~1ExAs

3-512

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAs. TEXAS 75265

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

Input SIgnal

I

r------------

I

Rf

RI ~
--11cl-~~~IN~II-~
R

-=-

I
I

The input resistance at each gain setting is given in Figure 28.
The -3 dB frequency can be calculated using equation 1:
f
1
-3 dB - 23t C(R II RI)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while value of the resistor to
ground should be decreased. In addition, the order of the filter could be increased.

input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, C, and the input impedance of the amplifier, Z" form a
high-pass filter with the comer frequency determined in equation 2.

fC(hlghpaSs) =

2nil c i

(2)

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3-513

TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
Input capacitor, CI (continued)
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where ZI is 710 ill and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
C

I

=_1_
2ltZl fC

(3)

In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 J.LF. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
that the source dc level. Note that it is important to confirm the capacitor polarity in the application.

power supply decoupllng, Cs
The TPA0162 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 J.LF placed as close as possible to the device Voo lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 J.LF or greater placed near
the audio power amplifier is recommended.

midrail bypass capacitor, CBYP
The mid rail bypass capacitor, CSyp, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CSyp determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CSyp, values of 0.47 J.LF to 1 J.LF ceramic or tantalum low-ESR capacitors are recommended
for the best THD and noise performance.

output coupling capacitor, Cc
In the typical sing ie-supply SE configuration, an output coupling capacitor (Cd is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

(4)

fC(high)

~TEXAS

3-514

INSTRUMENTS
POST OFFICE BOX 655303 • OAUAS, TEXAS 75265

TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S249B-JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

output coupling capacitor, Cc (continued)
The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency comer higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 JlF is chosen and loads vary from 3 n,
4 n, 8 n, 32 n, 10 kn, and 47 kn. Table 2 summarizes the frequency response characteristics of each
configuration.

Table 1. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode
Cc

RL

Lowest Frequency

30

330J,lF

161 Hz

40

330J,lF

120 Hz
60Hz

ao

330J,lF

320

330J,lF

15Hz

10,0000

330J,lF

0.05 Hz

47,0000

330J,lF

0.01 Hz

As Table 1 indicates, most of the bass response is attenuated into a 4-n load, an 8-n load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

bridged-tied load versus single-ended mode
Figure 32 shows a class-AB audio power amplifier (APA) in a BTL configuration. The TPA0162 BTL amplifier
consists of two class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 5).
V

_ VO(PP)
(rms) -

(5)

2/2

2
V(rms)

-R"L

Power -

~TEXAS

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3-515

TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load versus single-ended mode (continued)
VDD

oJ' ;

J'!

RL

vO(PP)

2x vO(PP)

Figure 32. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an a-n speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 33. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 ~F to 1000 ~F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.

~=

~

1
21tR L CC

For example, a 68-~F capacitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD

~dB~----~~====

fe

Figure 33. Single-Ended Configuration and Frequency Response

~TEXAS

3-516

INSTRUMENTS
POST OFFICE BOX 655303 • OALLAS. TEXAS 75265

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load versus single-ended mode (continued)
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor section.

single-ended operation
In SE mode (see Figure 32 and Figure 33), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SElBTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 1 VN.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
VOO' The internal voltage drop multiplied by the RMS value ofthe supply current,loOrms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understOOd (see Figure 34).
100

.'/
-~-

V(LRMS)

IOO(avg)

Figure 34. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

P

Efficiency of a BTL amplifier =

~

(7)

SUP
Where:
vLrms2

PL

= ~'

Vp

andV LRMS

= .f2'

Vp 2
therefore, PL = 2RL

*I

V
RP sin(t) dt =
o
L
:rt

and P SUP = VOolooavg and

looavg =

*

V:It
x RP [cos(t)] 0
L·

2Vp

= itA
L

~TEXAS

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POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-517

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS2498 - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

BTL amplifier efficiency (continued)
Therefore,

substituting PL a:nd Psup into equation 7,

V 2
P
2 RL

Efficiency of a BTL amplifier

2VOO Vp
1t RL

Where:

J2

Vp

P L RL

Therefore,

(8)
flBTL
PL = Power delivered to load
Psup Power drawn from power supply
VLRMS RMS voltage on BTL load
RL Load resistance
Vp = Peak voltage on BTL load
looavg = Average current drawn from the power supply
Voo = Power supply voltage
llBTL = Efficiency of a BTL amplifier

=

=
=

Table 2 employs equation 4 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is I~ss than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.

Table 2. Efficiency vs Output Power in 5-V 8-0 BTL Systems
OUTPUT POWER

EFFICIENCY
(%)

PEAK VOLTAGE
(V)

INTERNAL DISSIPATION

(W)

0.25

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.4rt

0.53

(W)

t High peak voltages cause the THD to incre.ase.
A final point to remember about class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

-!11 TEXAS

INSTRUMENTS
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TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature the internal
dissipated power at the average output power level must be used. From the TPA0162 data sheet, one can see
that when the TPA0162 is operating from a 5-V supply into a 3-n speaker that 4 W peaks are available.
Converting watts to dB:

= 10Log

P dB

(:w)
ref

=

10L09(~~) = 6 dB

(9)

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6
6
6
6
6

dB - 15 dB = -9 dB (15 dB crest factor)
dB -12 dB = -6 dB (12 dB crest factor)
dB - 9 dB = -3 dB (9 dB crest factor)
dB - 6 dB 0 dB (6 dB crest factor)
dB - 3 dB 3 dB (3 dB crest factor)

=
=

Converting dB back into watts:

P .

10 PdB /10 x P

W

ref

(10)

63 mW (18 dB crest factor)
125 mW (15 dB crest factor)
250 mW (9 dB crest factor)

=

500 mW (6 dB crest factor)
1000 mW (3 dB crest factor)
2000 mW (15 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-n system, the internal dissipation in the TPA0162 and
maximum ambient temperatures is shown in Table 3.
Table 3. TPA0162 Power Rating, 5-V, 3-a, Stereo
AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

4

2W(3dB)

1.7

4

1000 mW (S dB)

1.S

-3°C
SoC

4

500mW(9dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

4

63 mW (18 dB)

O.S

78°C
9Soe

PEAK OUTPUT POWER
(W)

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3-519

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
Sl0S249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
crest factor and thermal considerations (continued)
Table 4. TPA0162 Power Rating, 5-V, &-0., Stereo
PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

2.5W
2.5W
2.5W
2.5W

1250 mW (3 dB crest factor)
1000 mW (4 dB crest factor)
500 mW (7 dB crest factor)
250 mW (10 dB crest factor)

0.55
0.62
0.59
0.53

100°C
94°C
97°C
102°C

The maximum dissipated power, POmax , is reached at a much lower output power level for an 8 a load than for
a 3 load. As a result, this simple formula for calculating POmax may be used for an 8 application:

a

a

2VbD

PDmax = :n: 2R

(11)

L

a

However, in the case of a 3 load, the POmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 a load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table on page 4. Converting this to 0JA:

e JA

1
= _1_ = 450C/W
= Derating Factor
0.022

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given 0JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0162 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
TA Max = T J Max - 9 JA Po

(13)

= 150 - 45(0.6 x 2) = 96°C (15 dB crest factor)

NOTE:
Internal dissipation of 0.6 W Is estimated for a 2-W system with 15 dB crest factor per channel.

Tables 3 and 4 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0162 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 3 and 4 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-0 speakers dramatically increases the thermal performance by increasing amplifier
efficiency.

~TEXAS

3-520

INSTRUMENTS
POST OFFICE BOX 855303 • DAU..AS, TEXAS 75285

TPA0162
2-W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SLOS249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION

SE/BTL operation
The ability of the TPA0162 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0162, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When
SElBTL is held low, the amplifier is on and the TPA0162 is in the BTL mode. When SElBTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0162 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SElBTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 35.

20

RHPIN

23

RLINEIN

8

ROUT+

21

ROUT-

16

RIN

VDD~~~:
1k.O

_

100kn
SElBTL

15 100 kn

~

n

==t---=---'\I\I~

Figure 35. TPA0162 Resistor Divider Network Circuit
Using a readily available 1/S-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-knt1-kn divider pulls the SElBTL input low. When a plug is inserted, the 1-kn
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the OUT-amplifier is
shut-down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (Co) into the headphone jack.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-521

TPA0162
2·WSTEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S249B - JUNE 1999 - REVISED MARCH 2000

APPLICATION INFORMATION
PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is normally activated automatically, but may be selected
manually by pulling PCB ENABLE high. When the PC BEEP input is active, both of the LlNEIN and HPIN inputs
are deselected and both the left and right channels are driven in BTL mode with the Signal from PC BEEP. The
gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is independent of the volume setting. When
the PC BEEP input is deselected, the amplifier will return to the previous operating mode and volume setting.
Furthermore, if the amplifier is in shutdown mode, activating PC BEEP will take the device out of shutdown and
output the PC BEEP Signal, then return the amplifier to shutdown mode.
When PCB ENABLE is held low, the amplifier will automatically switch to PC BEEP mode after detecting a valid
signal at the PC BEEP input. The preferred input signal is a square wave or pulse train with an amplitude of 1
Vpp or greater. To be a accurately detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall
times of less than 0.1 !J.S and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier
will return to its previous operating mode and volume setting.
When PCB ENABLE is held high, PC BEEP is selected and the LlNEIN and HPIN inputs are deactivated
regardless of the input signal. PCB ENABLE has an internal 100 k.Q pulldown resistor and will trip at
approximately Vool2.
If it is desired to ac couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
the following equation:

C

>
1
PCB - 21t fpCB (100 kQ)

(14)

The PC BEEP input can also be dc coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.

~TEXAS

INSTRUMENTS

3-522

POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

TPA0162
2·W STEREO AUDIO POWER AMPLIFIER
WITH DIGITAL VOLUME CONTROL
SL0S249B - JUNE 1999 - REVISED MARCH 2000

APPliCATiON iNFORiViATiON

Input MUX operation
Right
Headphone
Input
Signal

CIRHP
0.47 I1F

----j

R
CIRLINE
0.4711F

23

MUX

RLINEIN

RlghtLlne~

Input
Signal

ROUT+

21

ROUT-

16

---;

8

RIN

T
Figure 36. TPA0162 Example Input MUX Circuit

Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SE/BTL operation section for a description of the headphone jack control circuit.

shutdown modes
The TPA0162 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, 100 = 150 !lA. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.
Table 5. Shutdown and Mute Mode Functions
AMPLIFIER STATE

INPUTSt
SElBTL

SHUTDOWN

INPUT

Low

High

Line

OUTPUT
BTL

X

Low

X

Mute

High

High

HP

SE

t Inputs should never be left unconnected.
X

=do not care

~1ExAs

INSTRUMENTS
POST OFFICE SOX 655303 • DALLAS, TEXAS 75265

3-523

3-524

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

• Integrated Depop Circuitry
• High Power with PC Power Supply
- 2 W/Ch at 5 V into a 3-0 Load
- 800 mW/Ch at 3 V
• Fully Specified for Use With 3-0 Loads
• Ultra-Low Distortion
- 0.05% THD+N at 2 Wand 3-0 Load
• Bridge-Tied Load (BTL) or Single-Ended
(SE) Modes
• Stereo Input MUX
• Surface-Mount Power Package
24-Pin TSSOP PowerPADTM
• Shutdown Control ••• 100

=5 ~A

PWPPACKAGE
(TOP VIEW)
GNDIHS
TJ
LOUT+
LLiNEIN
LHPIN
LBYPASS
LVDD
SHUTDOWN
MUTE OUT
LOUTMUTE IN
GND/HS

7

24
23
22
21
20
19
18

8
9
10
11
12

16
15
14
13

10
2
3
4
5
6

17

GND/HS
NC
ROUT+
RLiNEIN
RHPIN
RBYPASS
RVD D
NC
HP/LINE
ROUTSElBTL
GND/HS

RIR

21

--1
CIR

NC

RLiNEIN

20 RHPIN
19 RBYPASS

CB
System
Control

T-=-

-=-

11

MUTE IN

8

SHUTDOWN

Bias, Mute,
Shutdown,
andSE/BTl
MUXControl

6

lBYPASS

-=-

5

lHPIN

4

lLiNEIN

9 MUTE OUT

NC

--1

Ril

Left
MUX

Cil
CFl

4.

~

RFl

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments InCOrporated.

~~~~1:.=-~co:::"~::,c~

standard warranty. Production _109 doos not .........11y Include
II!IIlng of an pnmeIers.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3-525

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

description
The TPA0202 is a stereo audio power amplifier in a 24-pin TSSOP thermal package capable of delivering
greater than 2 W of continuous RMS power per channel into 3-0 loads. The TPA0202 simplifies design and frees
up board space for other features. Full power distortion levels of less than 0.1 % THD+N from a 5-V supply are
typical. Low-voltage applications are also well served by the TPA0202 providing 800-mW per channel into 3-0
loads with a 3.3-V supply voltage.
The TPA0202 has integrated depop circuitry that virtually eliminates transients that cause noise in the speakers
during power up and when using the mute and shutdown modes.
Amplifier gain is externally configured by means of two resistors per input channel and does not require external
compensation for settings of 2 to 20 in BTL mode (1 to 10 in SE mode). An internal input MUX allows two sets
of stereo inputs to the amplifier. In notebook applications, where internal speakers are driven as BTL and the
line (often headphone drive) outputs are required to be SE, the TPA0202 automatically switches into SE mode
when the SElBTL input is activated. Using the TPA0202 to drive line outputs up to 700 mW/channel into external
3-0 loads is ideal for small non-powered extemal speakers in portable multimedia systems. The TPA0202 also
features a shutdown function for power sensitive applications, holding the supply current at 51JA.
The PowerPAD packaget (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°CIW are readily realized in multilayer PCB
applications. This allows the TPA0202 to operate at full power into 3-0 loads at ambienttemperature of up to
85°C with 300 CFM of forced-air cooling. Into 8-0 loads, the operating ambient temperature increases to 100°C.
AVAILABLE OPTIONS
PACKAGE
TA

TSSOA

-40°C to 85°C

TPA0202PWP

(PWP)
:(: The PWP packages are available taped and reeled. To order a taped
and reeled part, add the suffix R (e.g., TPA0202PWPR).

t See Texas Instruments document, PowerPAD Thermally' Enhanced Package Application Report (Uterature Number SLMA002) for more
information on the PowerPAD package.

~1ExAs

3-526

INSTRUMENTS
POST OFFICE BOX 66530G • DAllAS, TEXAS 75265

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

Terminai Funciions
TERMINAL
NAME

NO.

GNOIHS

1,12,
13,24

HP/LINE

16

110

DESCRIPTION
Ground connection for circuitry, directly connected to thermal pad

I

Input MUX control input, hold high to select LHP IN or RHP IN (5, 20), hold low to select LLINE IN or
RLiNE IN (4, 21)

LBYPASS

6

LHPIN

5

I

Tap to voltage divider for left channel intemal mid-supply bias
Left channel headphone input, selected when HPILINE terminal (16) is held high

LLiNE IN

4

I

Left channel line input, selected when HP/LINE terminal (16) Is held low

LOUT+

3

0

Left channel + output in BTL mode, + output in SE mode

LOUT-

10

0

Left channel - output in BTL mode, high-impedance state in SE mode

LVOO
MUTE IN

7

I

Supply voltage input for left channel and for primary bias circuits

11

I

Mute all amplifiers, hold low for normal operation, hold high to mute

9

0

Follows MUTE IN terminal (11), provides buffered output

MUTE OUT
NC

No intemal connection

17,23

RBYPASS

19

Tap to voltage divider for right channel intemal mid-supply bias

RHPIN

20

I

Right channel headphone input, selected when HPILINE terminal (16) is held high

RLiNEIN

21

I

Right channel line input, selected when HP/LINE terminal (16) is held low

ROUT+

22

0

Right channel + output in BTL mode, + output in SE mode

ROUT-

15

0

RVOO
SElBTL

18

I

Supply voltage input for right channel

14

I

Hold low for BTL mode, hold high for SE mode

SHUTDOWN

8

I

TJ

2

0

Right channel- output in BTL mode, high impedance state in SE mode

Places entire IC in shutdown mode when held high, 100 = 511A
Sources a current proportional to the junction temperature. This terminal should be left unconnected
during normal operation. For more information, see the junction temperature measurement section of
this document.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-527

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS20SA - FEBRUARY 1998 - REVISED MARCH 2000

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, T J .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ........... . . . . . . . . . . . . . . . . . . .. 260°C

t

Stresses beyond those listed under,"absolute maximum ratings· may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

DERATING FACTOR
21.8mW/oC

2.7W

1.7W

1.4W

:I: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SlMA002), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
MIN

NOM

MAX

3

5

5.5

Supply Voltage, VOO

Operating free-air temperature, TA

Common mode input voltage, VICM

dc electrical characteristics, TA

VOO=5V,
250 mW/ch average power,

4-n stereo BTL drive,
with proper PCB design

-40

85

VOO=5V,
2 W/ch average power,

3-n stereo BTL drive,
with proper PCB design
and 300 CFM forced-air
cooling

-40

85

1.25

4.5

VOO=3.3V

1.25

2.7

V

=25°C
TYpt

MAX

Stereo BTL

19

25

mA

StereoSE

9

15

mA

Mono BTL

9

15

mA

MonoSE

3

10

mA

Stereo BTL

13

20

mA

StereoSE

5

10

mA

Mono BTL

5

10

mA

MonoSE

3

6

rnA

5

25

mV

TEST CONDITIONS

VOO=5V
Supply current

VOO=3.3V

VOO

Output offset voltage (measured differentially)

VOO=5V,

IOO(MUTE)

Supply current in mute mode

VOO=5V

1.5

IOO(SO)

100 in shutdown

VOO=5V

5

Gain =2,

NOTE 1: At 3 V < VOO < 5 V the dc output voltage is approximately VOot2.

~1ExAs

INSTRUMENTS
3--528

V

°C

VOO=5V

PARAMETER

100

UNIT

POST OFFICE BOX 655303 • OALLAS. TEXAS 75265

See Note 1

UNIT

mA
15

IlA

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SL0S205A - FEBRUARY 1998 - REVISED MARCH 2000

ac operating characteristics, VDD = 5 V, fA = 25u C, til = 3 n (uniess oiherwise noied)
PARAMETER
Po

Output power (each channel) see
Note 2

THD+N

Total harmonic distortion plus noise

BaM

TEST CONDITIONS

TYP

THD = 0.2%,

BTL,

See Figure 3

2

THD= 1%,

BTL,

See Figure 3

2.2

MAX

UNIT
W

Po = 2W,

1= 20-20 kHz,

See FigureS

200

m%

VI=l V,

RL=10kn,

AV= 1 VN

100

m%

Maximum output power bandwidth

AV=10VN

THD20

kHz

Phase margin

RL=4Q,

Open Loop,

See Figure 43

85°

1= 1 kHz,

See Figure 37

80

1= 20- 20 kHz,

See Figure 37

60

1= 1 kHz,

See Figure 39

Supply ripple rejection ratio
Mute attenuation

dB

85

dB

85

dB

LinelHP input separation

100

dB

BTL attenuation in SE mode

100

dB

2

MO

Channel-to-channel output separation

ZI

Input impedance
Signal-to-noise ratio

Po= 500 mW,

Vn

Output noise voltage

See Figure 35

BTL

95

dB

21

I1V(rms)

NOTE 2: Output power is measured at the output terminals 01 the IC at 1 kHz.

ac operating characteristics, VDD = 3.3 V, TA = 25°C, Rl =3 Q
PARAMETER
Po

Output power (each channel) see
Note 2

THD+N

Total harmonic distortion plus noise

BOM

TEST CONDITIONS

TYP

THD = 0.2%,

BTL,

See Figure 10

800

THD= 1%,

BTL,

See Figure 10

900

MAX

UNIT
mW

Po =8oomW,

1=20-20 kHz,

See Figure 11

350

m%

VI=l V,

RL=10kn,

AV= 1 VN

200

m%

Maximum output power bandwidth

AV= 10VN

THD20

kHz

Phase margin

RL=40,

Open Loop,

See Figure 44

85°

Supply ripple rejection ratio

1=1 kHz,

See Figure 37

70

1 = 20-20 kHz,

See Figure 37

55

f= 1 kHz,

See Figure 40

Mute attenuation

dB

85

dB

85

dB

LineIHP input separation

100

dB

BTL attenuation in SE mode

100

dB

2

MQ

Channel-to-channel output separation

ZI

Input impedance
Signal-to-noise ratiO

Po =500mW,

Vn

Output noise voltage

See Figure 37

BTL

95

dB

21

I1V(rms)

NOTE 2: Output power is measured at the output terminals 01 the IC at 1 kHz.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-529

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1996 - REVISED MARCH 2000

PARAMETER MEASUREMENT INFORMATION

-1 ~VV\,-e--+--t
4.711F

II
...L CB

SE/BTL - f - - ,

-=-

HP/LINE

Figure 1. BTL Test Circuit

Co

~
Voo
SE/BTL

-=-

HP/LINE

Figure 2. SE Test Circuit

~1ExAs

INSTRUMENTS
3--530

POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

RL=3Q,8Q,or32n

1

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTiCS
Table of Graphs
FIGURE

4,5,7,8,11,12,14,15,17,18,20,
21, 23, 24, 26, 27, 29, 3032, 33
3,6,9,10,13,16,19,22,25,28,31,
34
35,36
37,38
39-42

vs Frequency
THO+N

Total harmonic distortion plus noise

Vn

Output noise voltage

vs Frequency

Supply ripple rejection ratio

vs Frequency

Crosstalk

vs Frequency

vs Output power

Open loop response

vs Frequency

43,44

Closed loop response

vs Frequency

45,48
49
50,51
52,53

100

Supply current

vs Supply voltage

Po

Output power

vs Supply voltage
vs Load resistance

Po

Power dissipation

vs Output power

TOTAL HARMONIC DISTORTION PLUS NOISE

10
~
I

••z+
0

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY
10

I~ VOO=5V
I-f=lkHz
I- BTL

-L
-L

L

L

1

II

c

~

..

lL
RL=3Q

~

VOO=5V
PO=1.5W
RL=4Q
BTL

I

.~0
z

+
c

IIIIIII

0

'E

.s

0

is

..

I

'2

U

~
J:

0

S

AV=-10VN(RL=3,~PO=2W)

AV=-10VN':\

1"'1

0

RL=8Q
0.1

~ 1=

"Iii
is

I

.!!
c

~
J:

54-57

0.1

r--....

S

~

....

I

Z

+
Q

Ut ~'"

I

AV=-2VN -

+

J:

I 1111Ilil-

J:

I-

I-

o

V

~

Z

10"'"

Q

0.01

~

Av=-20VN~

0.25 0.5 0.75

1

1.25 1.5 1.75

2

2.25 2.5

0.01
20

Po - Output Power - W

IIIIIII
100

1k

10 k 20 k

f - Frequency - Hz

Figure 3

Figure 4

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-531

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SL0S205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY
1/1.

10

1/1.

VOO=5V
RL=4Cl
AV=-2VN
BTL

I

I
+
r::

VOO=5V
RL=3Cl
BTL

I

+

......... ......

g

!
i

PO=1.5W

PO=2W, RL=3Cl
I

I

Po =0.75

ij

0.1

/~

I I I III

~

W\

~
I

I ~O,i~~,fW

Z

eli

i!:

10

I

0

I

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

100

20

1k
f - Frequency - Hz

f=2OkHz

is

1'"

~

I

-... ....

0.1

I

I

10k 20k

0.01
0.01

111111

I

0.1
Po - Output Power - W

10

10

1/1.

VOO=5V
RL=SCl
AV =-2 VN
BTL

=

'0

z

VOO=5V
PO=1W
RL=SCl
BTL

I

~

Z

+
r::

+
r::

~

~

i
W~ I

.2
r::

0

0

Po = 0.5
0.1

~

.,

PO=1W

I

Z

......

eli

II
II

0.01
20

100

I

II

II

V

r--

".....

AV=-10VN

AV =-20 VN

;;3
/

~
I

AV =-2 VN

Z

Po =0.25 W

_

I I 1111111

1k
f - Frequency - Hz

0.1

eli

i!:

10k 20k

11111111

0.01

20

Figure 7

100

1k
f - Frequency - Hz

FigureS

~1ExAs

3-532

I

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

10

I

i!:

"111111

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

i
~
I

-

1=1 kHz

~

FigureS

1/1.

f~~Hz

~

i!:

LIIIIII

0.01

i

INSTRUMENTS
POST OFACE BOX 115S303 • DAllAS, TEXAS 75265

II il J

10k 20k

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPiCAL CHARACTERiSTiCS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10

'#.

VOO=5V
RL=80
AV=-2VN
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10

r- VOO=3.3V

..
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i= f= 1 kHz

CD

r- BTL

(5
Z

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+
c

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f=2OkHz

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10

o

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
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vs

FREQUENCY

FREQUENCY

'#.

VOO=3.3V
RL=40
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..
CD

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z

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,
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C

~

0

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10

I

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c

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TOTAL HARMONIC DISTORTION PLUS NOISE

vs
VOO=3.3V
Po = 0.75 W
RL=40
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0
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1

Figure 10

10

CD

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I

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Figure 9

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I

I

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~

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Po = 800 mW
(RL=30)
-

c

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0

..
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/...011~

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20

100

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~
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+

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-

10 k

:r
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20

Figure 11

100

1k
f - Frequency - Hz

10 k 20k

Figure 12

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-533

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SL0S205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

~
I

vs

OUTPUT POWER

FREQUENCY

10

~

Yoo =3.3Y
RL=30
Ay =-2 YN
BTL

Iz

+
c

YOO=3.3Y
PO=0.4W
RL=SO
BTL

j
+

0

f= 20 kHz

'E
0

j

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Q

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10

I

c

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vs

0.1

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f=20Hz

"-

::£:

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I

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low.

0.1

0::£:

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0.01

0.01
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20

2

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

FREQUENCY

OUTPUT POWER
~

~ YOO=3.3Y
I- RL=SO
I- Ay=-2YN
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10
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RL=SO
Ay =-2 YN
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I
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0

v

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1k

vs
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~
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0.1

r--...

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Figure 15

f = 1 kHz

1111

-.l

f=20Hz

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I

0.1
Po - Output Power - W

Figure 16

~TEXAS

3-534

100

Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

100

HUll 11
f - Frequency - Hz

Figure 13

20

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1IIIlii

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/

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I

+
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i', .....

os

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

10

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000
_ . _ . _ .. I

_ . . . . . . . #Ilio. . . . . . . . IIf"tIo . . .

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at.

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vs

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SE

at.

10
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I

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:
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vs

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V

1

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100

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10k 20k

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f - Frequency - Hz

Figure 17

Figure 18
TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

at.

VOO=5V
RL=40
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SE

r1

L.
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0

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10

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10

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CD

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0.01
20

0.01
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100

1k

1

10 k 20 k

f - Frequency - Hz

Figure 19

Figure 20

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-535

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

#.

vs

FREQUENCY

OUTPUT POWER

10
YOO=5Y
RL=80

I

~Z

I

Z

+
c

YOO=5 Y
RL=80
Ay=-2YN

r-

SE

~

.s

i

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f=20kHz

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0

E

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r-

~

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J:

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+
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vs

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J:

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100

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C
J:
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0

..

E
!

Po =0.25 W

;2I

c

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l1l
Figure 22

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

10

I
CD
III

'0

vs

vs

FREQUENCY

FREQUENCY

C YOO=5Y
;:: PO=0.075W
'- RL= 32 0
r

Z

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SE

z

+
c

~

0

.s
is

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c

./

0

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110

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10

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100

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10 k 20 k

1k

0.01
20

100

1k

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f - Frequency - Hz

Figure 23

Figure 24

-!11 TEXAS

INSTRUMENTS
3-536

/'

0.1
0.01
Po - Output Power - W

Figure 21

#.

/

~

f = 100 Hz
0.01
0.001

10 k 20 k

1k

f= 1 kHz

......

POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

10 k 20 k

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPiCAL CHARACTERiSTiCS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

1ft.

10

I

3:
·0

vs

OUTPUT POWER

FREQUENCY

r= VOO=5V
I- RL=320
~

z

vs
1ft.

SE

"

~0

~

...

~

~0

..

"

z0

+
c

~

Voo = 3.3 V
PO=0.2W
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GI

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+
c

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Ii

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0.1

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I

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I

f=20Hz /

Z

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Q

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0.01
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+
Q

Yl f=1kHz
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III

....

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Z

I

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I-

0.01

0.01
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100

20

Figure 25

vs

FREQUENCY

OUTPUT POWER

10

=

Voo = 3.3 V
RL=40
SE

3:

"0

:
-

z

+

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//

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is

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100

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I

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II

f = 1 kHz

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~

PO=0.2W

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RL=40
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I

c

is

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10

'1fl.

+

c

~

10 k 20 k

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

I

~
~

1k
f - Frequency - Hz

Figure 26

TOTAL HARMONIC DISTORTION PLUS NOISE

..

V

~

{!.

1ft.

/

AV=-10V/v

~
po"

Z

,~O=O.05W

IIII
1k

Q

r-

I-

f = 100 Hz

+

:z:

II
10 k 20 k

0.01
0.001

f - Frequency - Hz

IlL

0.01

0.1

Po - Output Power - W

Figure 27

Figure 28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-537

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORnON PLUS NOISE

'at

vs

vs

FREQUENCY

FREQUENCY

10

'at

Voo = 3.3 V
PO=1oomW
RL=SO
SE

I

j
z0

+

+

s

0

i:

I

M
~0

j
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VOO=3.3V
RL=SO
SE

J

c

E
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10

I

AV=-10VN
0.1

r-

I

Z

0

j!!
20

~ ....

AV=-1 VN

t--

II

1k
f - Frequency - Hz

I

r-

~11I~50:nW

~

Z

0

r--'""

0.01
20

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

'at

I

Iz

o
j!!

10
VOO = 3.3 V
PO=30mW
RL=320
SE

I

I

j

10k 20k

Figure 30

'at

7z

1k

f - Frequency - Hz

TOTAL HARMONIC DISTORnON PLUS NOISE

+

................

I
I

~

100

Figure 29

+

L~

Po=25mW

~

j!!

10 k 20k

'/

I UIII l

0.1

I

I

100

E
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PO=1oomW
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11111

0.01

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f:j

c

0

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~

is

AV=-10VN

u

................

0.1

c0
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j
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I---P-Iod f = 100 Hz

Z
+
C

........ "'

1111111 I .
1IIILll-l0.1

/

AV=-5VN

I

AV=-1 VN
WI

.....

j!!

IIII

0.01
20
Po - output Power - W

Figure 31

100

1k
f - Frequency - Hz

Figure 32

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

.1

1.

I

Ij
10k 20k

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SL0S205A- FEBRUARY 1998 - REVISED MARCH 2000

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

'#.

10

.;

Iz

Z

+

c

c

~

ii

is

0.1

PO=20mW

0

~I

,
0.01

Z

~

.i
Q

.2
c

0.1

0

Po=30mW

i
:c
!

P'"

i
:c

I

..n

~ f=20kHz

8

~

0

'2

SE

+

t:0
u

Voo = 3.3 V
RL=32Q

I

SE

0

10

'#.

VOO=3.3V
RL=32Q

I

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

~ ~f=1kHz
~ F--!..:. 20 Hz

!

~

0.01

I

PO= 10mW

z+

+
Q

Q

:c

i!:

II IIII

I-

0.001
20

100

1k

0.001
0.001

10 k 20 k

f - Frequency - Hz

0.01

Figure 33

Figure 34

OUTPUT NOISE VOLTAGE
vs
FREQUENCY

OUTPUT NOISE VOLTAGE
vs
FREQUENCY

100

100
Voo=S"V
BW = 22 Hz to 22 kHz
RL=4Q

'ii'

.[

,

VOBTL

I

-

.[
>::j.

VOBTL

I

III

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Voo = 3.3 V
BW = 22 Hz to 22 kHz
RL=4Q

'ii'

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~

0.1

Po - Output Power - W

III
01

Vo+

II

10

Vo-

:;

Vo+

!l

r=::

~

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-

-

jI

10

.!!
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==

:;

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::I

0

0

I

I

C

c

>

>
1
20

1
100

1k

10 k 20 k

20

100

1k

f - Frequency - Hz

f - Frequency - Hz

Figure 35

Figure 36

10k 20k

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-539

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO

SUPPLY RIPPLE REJECTION RATIO

vs

vs

FREQUENCY

FREQUENCY

0
III

-10

I

-20

0
RL=4Q
CB=4.7!lF
BTL

'0
0

!c:
0

11
I

I

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f

III

'0
I
0

ia:

-30

c:

0

-40

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-60

.........:::

-60
VOO=3.3V
-70
-80

...

II :11[" ~"'"

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20

100

-30

I"'III=::::
-40

'--

-60

Q.

-60

a

-70

III

-80

20

1k

100

CROSSTALK

III

'0
I

...

Ie
0

-70
-80

vs

vs

FREQUENCY

FREQUENCY

VOO=5V
PO=1.5W
RL=4Q
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~

'r--

IIILeft
r-...

-90

-50 r-60

L

i'.

/'

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.....

Right to Left

./

'"

V
~

III

'0
I

...

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~

0

JL..M

-120
10k 20 k

" .....

~

-110

1k
f - Frequency - Hz

L
/

V

-90 r- Right to Left

-110

100

1' . . .

-80 ~

-100

20

VOO=3.3V
Po = 0.75 W
RL=4Q
BTL

-70

-100

-120

10k 20 k

Figure 38

-40

-60

./

f - Frequency - Hz

CROSSTALK

-

11·1

-100

10k 20 k·

1k

Figure 37

-50

i'-.l

VOO=3.3V

f - Frequency - Hz

-40

VOO=5V

r--""

-90

11I1111

-100

-20

ii:

Q.
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VOO=5~11

-90

l
t

RL=4Q
CB=4.7!lF
SE

-10

20

100

1k
f - Frequency - Hz

Figure 40

Figure 39

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAu.AS, TEXAS 75265

V
:--

10k 20 k

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

--_._ .. _.... _......................,..,..

• 'I"""'AL ",nAnA'" IlI:nlo) 1 1"'0)

CROSSTALK
-40
-50

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY
-40

_~~:::~w
VOO=5V

VOO=3.3V

~~:::~w

-50 r-

SE

-eo
III

'D
I

...

Ie
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-70

-eo

.....

-70 t'-.

III
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I

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-80

...

r--..r--.

..... t--

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Right to Left

-110
-120
20

~

.....
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~
Right to

III

Left to Right

......

r--.. ....

-100

.> ~

~

i"o..

-80

I"

Left to Right

-100

SE

>

Le~"""

-110

111111
100

1k

-120

10k 20k

::::::~

11I1111
20

1k

100

f - Frequency - Hz

f - Frequency - Hz

Figure 41

Figure 42

10 k 20k

OPEN LOOP RESPONSE
100
VOO=5V
RL=4n

80
60
III

'D
I

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IllllL
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!'

40

Til

r--.

c

'li

CJ

180°

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20

"

90°'

0°

r-...

...

0 I-20 r-40 I 0.01

0.1

10

100

-

1000

_180°

10000

f - Frequency - kHz

Figure 43

:ilTEXAS

INSTRUMENTS

POST OFACE BOX 655303 • DALLAS. TEXAS 75265

3-541

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SL0S205A- FEBRUARY 1998- REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OPEN LOOP RESPONSE
80

r-.

60

VOO=3.3V
RL=4Cl
BTL

I-

rJ

~

II

l-

r..

40

I-

'\

ID

'1:1
I

iCI

20

,

INIIII

0

~
-20
-40
0.01

0.1

10
100
f - Frequency - kHz

1000

-1SOO
10000

Figure 44
CLOSED LOOP RESPONSE
10
VOO=5V
AV =-2 VN
PO=1.5W
BTL

9

8

_45°

7
Gain
6

I

~

5

I

-900

-135°

J•
Do

4

Phase

3

_180°

~

2

_225°

o

20

100

1k

10k

l -2700
100k 200k

f - Frequency - Hz

Figure 45

I~TEXAS
NSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

CLOSED LOOP RESPONSE
10

9

Voo = 3.3 V
AV=-2VN
Po = 0.75W

8

BTL

7

/

Gain

6

1/

~

5
4

....

Phase

3
".

2

o

20

,

100

-270·
lOOk 200k

lk
10k
f - Frequency - Hz

Figure 46
CLOSED LOOP RESPONSE
0

1/

-1

/

-2

~a~J I

-3
III

-4

"

-6

CI

-8

I

~

I
Phase

-7
VOO=5V
AV=-l VN
PO=0.5W
SE

..... ~

-8
-9

11111

-10
20

100

111111111

lk
10k
f - Frequency - Hz

-270·
lOOk 200k

Figure 47

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TPA0202
2~W STEREO AUDIO POWER AMPLIFIER
SL0S205A - FEBRUARY 199B - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
CLOSED LOOP RESPONSE
0
-1

I

-2

b~i~ I

IL

_45°

II

-3
ID

_90°

-4

I

'C

I

-5

CJ

-6

c
'ii

-135°

J
Do.

Phase
-180°

-7

i'

-8

VDD= 3.3V
AV=-1 VN
PO=0.25W
SE

-9
-10
20

100

-

_225°

-

1111
I 11111111
-270°
10 k
100k 200k
1k
f - Frequency - Hz

Figure 48
SUPPLY CURRENT

OUTPUT POWER

vs

vs

SUPPLY VOLTAGE

SUPPLY VOLTAGE

30
THD+N

BTL

25

2.5

1

~

I

I

"E

I

~

Do.

~

!:i

D!

0

a.
a.

=1%

Each Channel +-+---t-~'+---l

2

1.5

t

I

I

,p

Q

E

0.5

O~

3

______

~

________

~

______

4
5
VDD - Supply Voltage - V

~

6

0
2.5

3

3.5

4

4.5

5

VDD - Supply Voltage - V

Figure 49

Figure 50

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5.5

6

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

OUTPUT POWER

OUTPUT POWER

vs

vs

SUPPLY VOLTAGE

LOAD RESISTANCE
3

THD+N = 1%
SE
Each Channel

2.5

0.8
~

~

I

i
0

RL=4n/
0.6

D.

'5
,e.
::s

0

I

I

/i
./

D.

'5

0.4

I

.....

/
/'

cP
0.2

V
o

3

2.5

,......., ~

/"

\

2

\\

1.5

~

RL=8n

0

V

-,

cP
0.5

-

3.5
4
4.5
5
VDD - Supply Voltage - V

o

5.5

~D=5V

...

I

RL=32n

THD+N = 1%
BTL
Each Channel

\

6

"

'",

. . . . . . r--

VDD=3.3V

o

4

--

.............

-

8
12
16
20
24
RL - Load Resistance - n

Figure 51

28

POWER DISSIPATION

vs

vs

LOAD RESISTANCE

OUTPUT POWER
1.8

I

0.6

'5

~

~

c
0

!..

0.4

\

\

I

cP
0.2

j

"'~
"'~

VDD=3.3V

o

I'

o

4

1.2

is

!--

0.8

0

VDD=5V

D.
I

--

c

/'

1.4

I

;\

D.

0

1.6

\

~

I

THD+N = 1%
SE
Each Channel

\

32

Figure 52

OUTPUT POWER

0.8

r--

0.6

D.

.-

I ...........-:RL=4Q

/1
1/

VRL ..

D

r- __ ..

~ R~=3Q

r-..

0.4

r-- to-

8
12
16
20
24
RL - Load Resistance - n

28

32

VDD=5V
BTL
Each Channel

0.2

o
o

0.5

1.5

Po - Output Power Figure 54

Figure 53

2

2.5

W

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TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
POWER DISSIPATION
vs
OUTPUT POWER

---........

0.8
""""""-RL=3n

0.7

==cI

0.6

0

:;

CI.

0.5

C

0.4

.,

'iii

~

0

Il.
I

0.3

I V
//
V
rt/ --..

RL=4n

RL=8n

C

Il.

I

/ V

----,

0.8

3:
I

0.6

RL=~

c

i

'iii

is

t

0.4

I

"'-

0.2

a.

........

I

C

a.

0.2

f(

VOO = 3.3 V
BTL

0.1

o
o

POWER DISSIPATION
va
OUTPUT POWER

Each Channel

o

0.25
0.5
0.75
Po - Output Power - W

V

RL =32n

~

o

--

--

RL=8n

V-

VOO=5V
SE
Each Channel

0.1

0.2

0.3

0.4

Po - Output Power - W

Figure 55

Figure 56
POWER DISSIPATION
vs
OUTPUT POWER

0.6..-----,----,.---r---...,-----,
VOO=3.3V
SE
Each Channel

==cI
o

Ic
J
I

rP

0.25
Po - Output Power - W

Figure 57

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0.6

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SL0S205A - FEBRUARY 1998 - REVISED MARCH 2000

THERMAL iNFORiviATiON
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 58)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface-mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Slde~(a)

Thermal
Pad

EndVl_(b)

Bottom VI_ (e)

Figure 58. Views of Thermally Enhanced PWP Package

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2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

bridged-tied load versus slngle-ended mode
Figure 59 shows a linear audio power amplifier (APA) in a BTL configuration. The TPA0202 BTL amplifier
consists of two linear amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration but initially consider power to the load. The differential drive to the speaker means
that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the voltage
swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power equation, where
voltage is squared, yields 4x the output power from the same supply rail and load impedance (see equation 1).
V (rms) =

VO(PP)
2./2
V(rms)

2

(1)

-RL

Power -

voo

V' ~

RL

J'!
rv ~

VO(PP)

2x VO(PP)

-VO(PP)

Figure 59. Bridge-Tied Load Configuration

In a typical computer sound channel operating at 5 V, bridging raises the power into an a-n speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 60. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 3311F to 1000 I1F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 2.
(2)

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2·W STEREO AUDIO POWER AMPLIFIER
SL0S205A- FEBRUARY 1998 - REVISED MARCH 2000

APPLiCATiOn inFOFiiviAiiON
bridged-tied load versus single-ended mode (continued)
For example, a 68-J.LF capacitor with an 8-0 speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

VOO

~dB~----~~====

Figure 60. Single-Ended Configuration and Frequency Response
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the thermal considerations
section.

BTL amplifier efficiency
Linear amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across the
output stage transistors. There are two components of the internal voltage drop. One is the headroom or dc
voltage drop that varies inversely to output power. The second component is due to the sinewave nature of the
output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from Voo.
The internal voltage drop multiplied by the RMS value of the supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS values of power in the load and in
the amplifier, the current and voltage waveform shapes must first be understood (see Figure 61).
'DO

,/

V(LRMS)

-~-

IOO(RMS)

Figure 61. Voltage and Current Waveforms for BTL Amplifiers

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2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
Efficiency

P

L
=P

(3)

SUP

Where:

:It (

Efficiency of a BTL Configuration =

:It

p R )1/2
...b....b

VP

2

(4)

2VOO
Table 1 employs equation 4 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-n loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 1. Efficiency Vs Output Power in 5-V 8-n BTL Systems
PEAK-TO-PEAK
VOLTAGE
(V)

INTERNAL
DISSIPATION

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.4?t

0.53

OUTPUT POWER
(W)

EFFICIENCY

0.25

(%)

(W)

t High peak voltages cause the THO to increase.

A final point to remember about linear amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 4, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

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2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPliCATiON iNFORiviATiON
For example, if the 5-V supply is replaced with a 3.3-V supply (TPA0202 has a maximum recommended VDD
of 5.5 V) in the calculations of Table 1, then efficiency at 0.5 W would rise from 44% to 67% and internal power
dissipation would fall from 0.62 W to 0.25 W at 5 V. Then for a stereo 0.5-W system from a 3.3-V supply, the
maximum draw would only be 1.5 W as compared to 2.24 W from 5 V. In other words, use the efficiency analysis
to choose the correct supply voltage and speaker impedance for the application.

selection of components
Figure 62 and Figure 63 are a schematic diagrams of a typical notebook computer application circuits.

CFR
5pF

t

~~

RFR
501en

~--------------------------~
RIR

----1~
CIR
111F

1° kQ

21

RLINEIN

ROUT+ 22

Right 1--_ _-1
NC ----'2=.=0+R-"H-"P-"IN'-'-_ __1 MUX
:
19

RBVPASS

-

II

ROUT- 15
I

RVDD 18

T
-::!:::-r

System
11 MUTE IN
Control --+---....:...:...t-=-='=.:.=-=-----1
9 MUTE OUT

I

8

SHUTDOWN

Blas,Mute,
Shutdown,
andSElBTL
MUXControl

_
SE/BTL 14 -

LBVPASS

-

RIL NC _-=5+L:::.H.:.:.P-"IN-=---_ __1
Left
+
L 10 len
----1~r-~~Y--,--4~=LL=IN~E=I~N-__1 MUX 1--_ ___1-

-

LOUT+

C

O'~I1F
(see Note A)~

.l
T
J

HP/LINE 16..

I-----~~~~~

LVDD 7

6

-:::'--- COU;-~I' 330 l1F
VDD
100 kQ

1 len

-do-=-

100kQ
0.111F

VDD

3

LOUT- 10
I

NOTE A. A O.1I1F ceramic capacitor should be placed as close as possible to the IC. Forfmering lower-frequency noise signals, a larger aluminum
electrolytic capacitor of 10 flF or greater should be placed near the audio power amplifier.
.

Figure 62. TPA0202 Minimum Configuration Application Circuit

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2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

CFRLlNE.
5pF
RFRHP
10kO

-q'
~
,
"

RFRLlNE
50kO

CI~LrE RIRLINE

~f JO kn

21

RLINEIN

.,

:--Right 1---_


--1tf------'\Nv--<___-=2=-0t...:..=:RH:..::P'-"IN~-_I_MUX
RIRHP
C
IRHP 10kn
1liF
CBR

~

:

~

ROUT+ 22

II

ROUT- 15

19 RBYPASS
..........
I,.---""-I-'~~""'--------'T
RV-- 18
;:::::; CoUTR
±
r--A./\tv--.....
~-'VI/\r---.:..:..:.J'u......
ul-'-"'-*----...-VDD
330J,lF

J:. ~~~ ? i
T
_ (~NoteB~ 10011n 1 lin ~

J.

0.1 liFT

-=-

System _ _ _ _-'1.;...1rM",Uc..:.T=E..:.:IN""""--I Bias, Mute,
Control
a~~u~~:i.
9 MUTE OUT
See Note A [
8 SHUTDOWN MUX Control

SE/BTL 14 _

or

100 lin
HP/LINE 16

::E 0•1 J,lF
LVDD
6

LBYPASS

5

LHPIN

7

T

-::!:::-

T-=-

~T

-=-

VDD
CSR
0.1 J.lF
(see Note B)

CBL
1liF _
CILHP
1liF

--1111

RILHP
10kO

-

Left
4 LLiNEIN
MUX
+
7~r~~~~===-~
r--~RILLINE
CILLlFNE 10 lin

h

\L

LOUT+ 3
LOUT- 10

II

11i

RFLHP
10 lin

D
"T

CFLLlNE;
5P

RFLLINE
50kn

NOTES: A. This connection is for ultra-low current in shutdown mode.
B. A 0.1 IiF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
aluminum electrolytic capacitor of 10 IiF or greater should be placed near the audio power amplifier.

Figure 63. TPA0202 Full Configuration Application Circuit

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TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SlOS205A - FEBRUARY 1998 - REVISED MARCH 2000

AppLiCATiON iNFORMATiON
gain setting resistors, RF and R,
The gain for each audio input of the TPA0202 is set by resistors RF and RI according to equation 5 for BTL mode.
BTL Gain

= -

2(~~)

(5)

BTL mode operation brings about the factor 2 in the gain equation due to the inverting amplifier mirroring the
voltage swing across the load. Given that the TPA0202 is a MOS amplifier, the input impedance is very high,
consequently input leakage currents are not generally a concern although noise in the circuit increases as the
value of RF increases. In addition, a certain range of RF values is required for proper start-up operation of the
amplifier. Taken together it is recommended that the effective impedance seen by the inverting node of the
amplifier be set between 5 k.Q and 20 kn. The effective impedance is calculated in equation 6.

R R
Effective Impedance = R F ~
F+ I

(6)

As an example consider an input resistance of 10 k.Q and a feedback resistor of 50 kn. The BTL gain of the
amplifier would be -10 and the effective impedance at the inverting terminal would be 8.3 kn, which is well within
the recommended range.
For high performance applications metal film resistors are recommended because they tend to have lower noise
levels than carbon resistors. For values of RF above 50 k.Q the amplifier tends to become unstable due to a pole
formed from RF and the inherent input capacitance of the MOS input structure. For this reason, a small
compensation capacitor of approximately 5 pF should be placed in parallel with RF when RF is greater than
50 kn. This, in effect, creates a low pass filter network with the cutoff frequency defined in equation 7.

4dB~====~~----(7)

fe(IOwpasS)

For example, if RF is 100 k.Q and Cf is 5 pF then fe is 318 kHz, which is well outside of the audio range.

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2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICAtiON INFORMATION
Input capacitor, C,
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, CI and RI form a high-pass filter with the comer frequency
determined in equation 8.

.
1
fc(highpass) = 2 3t RI C I

(8)

fe

The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where RI is 10 kn and the specification calls for a flat bass response down to 40 Hz.
Equation 8 is reconfigured as equation 9.
C =_1_
I
2nRl f C

(9)

In this example, CI is 0.40 I1F so one would likely choose a value in the range of 0.47 I1F to 1 I1F. A further
consideration for this capacitor is the leakage path from the input source through the input network (RI' CI) and
the feedback resistor (RF) to the load. This leakage current creates a de offset voltage at the inputto the amplifier
that reduces useful headroom, especially in high gain applications. Forthis reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor.
should face the amplifier input inmost applications as the dc level there is held at Vool2, which is likely higher
that the source de level. Please note that it is important to confirm the capacitor polarity in the application.
power supply decoupllng, Cs
The TPA0202 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 I1F placed as close as possible to the device Voo lead works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 I1F or greater placed near
the audio power amplifier is recommended.

~1ExAs

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TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS20SA - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
midrall bypass capacitor, CB
The midrail bypass capacitor, Ca, is the most critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, Ca determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N. The capacitor is fed from a 100-1<0 source inside the amplifier. To keep the start-up pop as low as
possible, the relationship shown in equation 10 should be maintained.

1
(C a x 100

1

<

kn) -

C,(R,

+

(10)

R F)

As an example, consider a circuit where CB is 1 IlF, C, is 0.22 IlF, RF is 50 kn, and R, is 10 k.O. Inserting these
values into the equation 10 we get 10 :::; 75, which satisfies the rule. Bypass capacitor, Ca, values of 0.1 IlF to
1 IlF ceramic or tantalum low-ESR capaCitors are recommended for the best THO and noise performance.
In Figure 63, the full feature configuration, two bypass capacitors are used. This provides the maximum
separation between right and left drive circuits. When absolute minimum cost and/or component space is
required, one bypass capacitor can be used as shown in Figure 62. It is critical that terminals 6 and 19 be tied
together in this configuration.

load considerations
Extremely low impedance loads (below 4 0) coupled with certain external component selections, board layouts,
and cabling can cause oscillations in the system. Using a Single air-cored inductor in series with the load
eliminates any spurious osciffations that might occur. An inductance of approximately 1 IlH has been shown to
eliminate such oscillations. There are no special considerations when using 4 0 and above loads with this
amplifier.

optimizing depop operation
Circuitry has been included in the TPA0202 to minimize the amount of popping heard at power-up and when
coming out of shutdown mode. Popping occurs whenever a voltage step is applied to the speaker. If high
impedances are used for the feedback and input resistors, it is possible for the input capacitor to drift downwards
from mid-rail during mute and shutdown. A high gain amplifier intensifies the problem as the small delta in
voltage is multiplied by the gain. So it is advantageous to use low-gain configurations, and to limit the size of
the gain-setting resistors. The time constant of the input coupling capaCitor (C,) and the gain-setting resistors
(R, and RF) needs to be shorter than the time constant formed by the bypass capacitor (Ca) and the output
impedance of the mid-rail generator, which is nominally 100 1<0 (see equation 10).
The effective output impedance of the mid-rail generator is actually greater than 100 1<0 due to a PNP transistor
clamping the input node (see Figure 64).

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2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

Voo
1001<0
BYPASS -

......-A!II'v-......-

..
100kO

Figure 64. PNP Transistor Clamping of BYPASS Terminal
The PNP transistor limits the voltage drop across the 50 kQ resistor by slewing the internal node slowly when
power is applied. At start-up, the xBYPASS capacitor is at O. The PNP is pulling the mid-point of the bias circuit
down, so the capacitor sees a lower effective voltage, and thus charges slower. This appears as a linear ramp
(while the PNP transistor is conducting), followed by the expected exponential ramp of an R-C circuit.

o

If the expression in equation 1 cannot be fulfilled or the small amount of pop is still unacceptable for the
application, then external circuitry must be added that can eliminate the pop heard during power up and while
transitioning out of mute or shutdown modes.
By holding the device in SE mode when the pop normally occurs, no pop can be heard through the
BTL-connected speakers (as the negative output is in a high impedance state when the amplifier is in SE mode).
From a hardware point of view, the easiest way to implement this is to drive the SElBTL terminal from the
general-purpose input-output(GPIO) in the system. If the SElBTL terminal is normally connected to a
headphone socket (as shown in Figure 65), then the GPIO signal must either be taken through an OR gate (see
Figure 65) or isolated with a diode (any signal diode) (see Figure 66).
VOO

Right
Channel

Rm1
1001<0

semTL~ ~1"t
From GPIO

Rm2
1001<0

Channel

-=-

Figure 65. Implementation with an OR Gate

~1ExAs

INSTRUMENTS
3-556

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
VDD

Right
Channel

Rm1
100kQ

Channel

From GPIO

-=-

Figure 66. Implementation with a Diode
The OA gate and diode isolate the GPIO terminal from the headphone switch. In these implementations, the
headphone switch has priority.
When the amplifier is in mute mode, the output stage continues to be biased. This causes the transition out of
mute mode to be very fast with only a short delay (from 100 ms to 500 ms). During power up or the transition
out of shutdown mode, a longer delay ( from 1 s to 2 s) is required. The exact delay time required is dependent
on the values of the external components used with the amplifier (see Figure 67).
System Control:
MUTE or SHUTDOWN
Delay

. . . . . - - -.... Ir--...;..~
Output of Delay Circuit
(Input to SE/BTL) _ _ _ _--'

Figure 67. Transition Delay Timing

single-ended operation
In SE mode (see Figure 59 and Figure 60), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 22 and 3).
In SE mode the gain is set by the AF and AI resistors and is shown in equation 11. Since the inverting amplifier
is not used to mirror the voltage swing on the load, the factor of 2, from equation 5, is not included.
SE Gain = -

(~~)

(11 )

The output coupling capacitor required in single-supply SE mode also places additional constraints on the
selection of other components in the amplifier circuit. The rules described earlier still hold with the addition of
the following relationship (see equation 12):
<_1_~_1_
1
ALC C
(C B x 251<0) - (CIA I)

(12)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-557

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter govemed by equation 14.

fC(high)

(14)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. ConSider the example where a Cc of 330 IlF is chosen and loads vary from 3 a,
4 n, 8 n, 32 n, 10 kn, to 47 ka. Table 2 summarizes the frequency response characteristics of each
configuration.
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics In SE Mode
RL

Cc

LOWEST FREQUENCY

30

330I1F

161 Hz

40

330 l1F

120Hz
60Hz

80

330 l1F

320

330l1F

15 Hz

10,0000

330l1F

0.05 Hz

47,0000

330I1F

0.01 Hz

As Table 2 indicates, most of the bass response is attenuated into a 4-a load, an 8-a load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

~TEXAS

3-558

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
SElBTL operation
The ability of the TPA0202 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0202, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 14)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 10 and 15). When
SElBTL is held low, the amplifier is on and the TPA0202 is in the BTL mode. When SElBTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0202 as an SE driver from LOUT+
and ROUT+ (terminals 3 and 22). 100 is reduced by approximately one-half in SE mode. Control of the SElBTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 68.

21

RLiNEIN

20

RHPIN

MUXt----i

ROUT+ 22

ROUT- 15
Rm3

1 lin

Bypass

7

VDD
Rm1
100 lin
SElBTL 14
'--_ _ _ _ _ _ _H:,::P:..:./L:::I:.:;NE=-t-'1""S.....

T_

~
O.1JtF

I
Left

Channa'

~. ~
v

-=

Figure 68. TPA0202 Resistor Divider Network Circuit

Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-kn/1-kn divider pulls the SElBTL input low. When a plug is inserted, the 1-kn
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the OUT-amplifier is
shutdown causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capaCitor (Co) into the headphone jack.
As shown in the full feature application (Figure 63), the input MUX control can be tied to the SE/BTL input. The
benefits of doing this are described in the following input MUX operation section.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-559

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SL0S205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
Input MUX operation
Working in concert with the SElBTL feature, the HPILINE MUX feature gives the audio designer the flexibility
of a multichip design in a single Ie (see Figure 69). The primary function of the MUX is to allow different gain
settings for BTL versus SE mode. Speakers typically require approximately a factor of 10 more gain for similar
volume listening levels as compared to headphones. To achieve headphone and speaker listening parity, the
resistor values would need to be set as follows:
SE Gain(HP) = _

(~(HP»)
I(HP)

(15)

If, for example RI(HP) = 10 kn and RF(HP) = 10 kn then SE Gain(HP) =-1
.
BTL Galn(LlNE)
= - 2 (RF(LlNE»)
-=R,-'----!.
I(LlNE)

(16)

If, for example RI(LlNE) = 10 kn and RF(LlNE) = 50 kn then BTL Gain(LlNE) = -10

RFRUNE
CIRUNE RIRUNE

~~~~-r~2~1~R=U=N~E~IN~
ROUT+
~~¥0~___~~~R~H~P~IN~

CiRHP RIRHP

22

ROUT- 15
RIght Channel

MID
VDD

SElBTL

T

O.1I1F

Left Channel

Figure 69. TPA0202 Example Input MUX Circuit
Another advantage of using the MUX feature is setting the gain of the headphone channel to -1. This provides
the optimum distortion performance into the headphones where clear sound is more important. Refer to the
SElBTL operation section for a description of the headphone jack control circuit.

~lExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SL0S205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION
mute and shutdown modes
The TPA0202 employs both a mute and a shutdown mode of operation designed to reduce supply current, 100,
to the absolute minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input
terminal should be held low during normal operation when the amplifier is in use. Pulling SHUTDOWN high
causes the outputs to mute and the amplifier to enter a low-current state, 100 = 511A. SHUTDOWN or MUTE
IN should never be left unconnected because amplifier operation would be unpredictable. Mute mode alone
reduces 100 to 1.5 mAo

Table 3. Shutdown and Mute Mode Functions
OUTPUT

INPUTSt

t

AMPLIFIER STATE

SE/BTL

HP/LINE

MUTE IN

SHUTDOWN

MUTE OUT

INPUT

Low

Low

Low

Low

Low

LlR Line

BTL

X
X

X
X

-

High

High

High

X
X

Mute

-

-

Low

High

Low

Low

Low

LlRHP

BTL

High

Low

Low

Low

Low

LIR Line

SE

High

High

Low

Low

Low

LlRHP

SE

OUTPUT

Mute

Inputs should never be left unconnected.
X do not care

=

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

s-y versus 3.3-Y operation
The TPA0202 operates over a supply range of 3 V to 5.5 V. This data sheet provides full specifications for 5-V
and 3.3-V operation, as these are considered to be the two most common standard voltages. There are no
special considerations for 3.3-V versus 5-V operation as far as supply bypassing, gain setting, or stability goes.
For 3.3-V operation, supply current is reduced from 19 mA (typical) to 13 mA (typical). The most important
consideration is that of output power. Each amplifier in TPA0202 can produce a maximum voltage swing of
Voo - 1 V. This means, for 3.3-V operation, clipping starts to occur when VO(PP) = 2.3 V as opposed to
VO(PP) = 4 V at 5 V. The reduced voltage swing subsequently reduces maximum output power into an 8-0 load
before distortion becomes significant.
Operation from 3.3-V supplies, as can be shown from the efficiency formula in equation 4, consumes
approximately two-thirds the supply power for a given output-power level than operation from 5-V supplies.
When the application demands less than 500 mW, 3.3-V operation should be strongly considered, especially
in battery-powered applications to improve the efficiency.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-561

TPA0202
2·W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

headroom and thermal considerations
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. From the TPA0202 data sheet, one can see that when the
TPA0202 is operating from a 5-V supply into a 3-n speaker that 2 W peaks are available. Converting watts to
dB:
10Log (P w )
P ref

(17)

10L09(f)
= 3.0 dB
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
3.0 dB - 15 dB = - 12 dB (15 dB headroom)
3.0 dB - 12 dB = - 9 dB (12 dB headroom)
3.0 dB - 9 dB = - 6 dB (9 dB headroom)
3.0 dB - 6 dB = - 3 dB (6 dB headroom)
3.0 dB - 3 dB= 0 dB (3 dB headroom)
Converting dB back into watts:

P w = 10PdB/10 x P ref

(18)

= 63 mW (15 dB headroom)
= 120 mW (12 dB headroom)
= 250 mW (9 dB headroom)
= 500 mW (6 dB headroom)
= 1000 mW (3 dB headroom)
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with 0 dB of
headroom, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 5-V, 3-n system, the internal dissipation in the TPA0202
and maximum ambient temperatures is shown in Table 4.

~TEXAS

INSTRUMENTS
3-562

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

headroom and thermal considerations (continued)
Table 4. TPA0202 Power Rating, 5-V,

3-n, Stereo

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

2

2W

1.7

-3°C

2

1000 mW (3 dB)

1.6

6°C

2

500mW (6 dB)

1.4

24°C

PEAK OUTPUT POWER
(W)

2

250 mW (9 dB)

1.1

51°C

2

120 mW (12 dB)

0.8

78°C

2

63 mW (15 dB)

0.6

96°C

DISSIPATION RATING TABLE
PACKAGE

TAS25°C

DERATING FACTOR

TA = 70°C

pwpt

2.7W

21.8mW/oC

1.7W

TA

=85°C

1.4W

22.1 mW/oC
1.4W
pwpt
2.8W
1.8W
tThls parameter IS measured with the recommended copper heat sink pattern on a Hayer PCB, 4 In2 5-ln x 5-ln PCB,
1 oz. copper, 2-in x 2-in coverage.
t This parameter is measured with the recommended copper heat sink pattern on an 8-layer PCB, 6.9 in 2 1.5-in x 2-in PCB,
1 oz. copper with layers 1, 2, 4, 5, 7, and 8 at 5% coverage (0.9 in2) and layers 3 and 6 at 100% coverage (6 in2).

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the 0 CFM
and 300 CFM data from the dissipation rating table, the derating factor for the PWP package with 6.9 in 2 of
copper area on a multilayer PCB is 22 mW/cC and 54 mW/cC respectively. Converting this to ElJA:

e JA =

1
Derating

(19)

For 0 CFM:

1
0.022
45°C/W
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given ElJA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0202 is
150 cC. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-563

TPA0202
2-W STEREO AUDIO POWER AMPLIFIER
SLOS205A - FEBRUARY 1998 - REVISED MARCH 2000

APPLICATION INFORMATION

headroom and thermal considerations (continued)
TA Max

T J Max - 9 JA Po

(20)

150 - 45(0.6 x 2) = 96°C (15 dB headroom, 0 CFM)
NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB headroom per channel.

Table 4 shows that for some applications no airflow is required to keep junction temperatures in the specified
range. The TPA0202 is designed with thermal protection that turns the device off when the junction temperature
surpasses 150a C to prevent damage to the IC. Table 4 was calculated for maximum listening volume without
distortion. When the output level is reduced the numbers in the table change significantly. Also, using 8-0
speakers dramatically increases the thermal performance by increasing amplifier efficiency.

junction temperature measurement
Characterizing a PCB layout with respect to thermal impedance is very difficult, as it is usually impossible to
know the junction temperature of the IC in question. The TPA0202 terminal 2 (TJ) sources a current proportional
to the junction temperature. The circuit internal to TJ is shown in Figure 70.

voo
R

R
5R
TJ - - - ' \ I I I \ r - - + - - - - l

Figure 70. T J Terminal Internal Circuit
Connect an ammeter between TJ and ground to measure the current. As the resistors have a tolerance of±20%,
this measurement must be calibrated on each device. The intent ofthis function is in characterization ofthe PCB
and end equipment and not a real-time measurement of temperature. Typically a 25°C reading is -120 J.IA for
a 3.3-V supply and -135 J.IA for a 5-V supply. The slope is approximately 0.25 fJAf'Cfor both VOO = 3.3 V and
Voo = 5 V. To reduce quiescent current, do not ground TJ in normal operation. It can be connected ,to Voo or
left floating as it has a resistor connected across the base-emitter junction.

~TEXAS

3-564

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

TPA0212
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
PWPPACKAGE
(TOP VIEW)

• Compatible With PC 99 Desktop Line-Out
Into 10-1<0 Load
• Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
• 2-W/Ch Output Power Into 3-0 Load

•
•
•
•
•
•
•

Input MUX Select Terminal
PC-Beep Input
Depop Circuitry
Stereo Input MUX
Fully Differential Input
Low Supply Current and Shutdown Current
Surface-Mount Power Packaging
24-Pln TSSOP PowerPADTM

GND
GAINO
GAIN1
LOUT+
LLINEIN
LHPIN
PVoo
RIN
LOUTLIN
BYPASS
GND

10
2
3
4
5
6
7

8
9
10
11
12

24
23
22

21
20
19
18
17
16
15
14
13

GND
RLINEIN
SHUTDOWN
ROUT+
RHPIN
Voo
PVoo
HP/LINE
ROUTSElBTL
PC-BEEP
GND

description
The TPA0212 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-0 loads. This device minimizes the number of
external components needed, simplifying the deSign, and freeing up board space for other features. When
driving 1 W into 8-0 speakers, the TPA0212 has less than 0.8% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
Amplifier gain is internally configured and controlled by way of two terminals (GAl NO and GAIN1). BTL gain
settings of 2,6, 12, and 24 VN are provided, while SE gain is always configured as 1 VN for headphone drive.
An internal input MUX allows two sets of stereo inputs to the amplifier. The HP/LINE terminal allows the user
to select which MUX input is active regardless of whether the amplifier is in SE or BTL mode. In notebook
applications, where internal speakers are driven as BTL and the line outputs (often headphone drive) are
required to be SE, the TPA0212 automatically switches into SE mode when the SElBTL input is activated, and
this reduces the gain to 1 VN.
The TPA0212 consumes only 6 mA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to less than 150 IJA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°CIW are readily realized in multilayer PCB
applications. This allows the TPA0212 to operate at full power into 8-0 loads at an ambient1emperature of 85°C.

A\..

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

Copyright © 1999. Texas Instruments Incorporated

3-565

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

functional block diagram
RHPIN----I
RLiNEIN - - - - I

>--+-------

ROUT+

>--+--1------

ROUT-

RIN - - - - - - - - - - - - - 1 -..

PC·BEEP

--1. ._:

e_c-_
ep---,
Power
Management

GAINO
GAIN1
SElBTL

PVDD
VDD
BYPASS
SHUTDOWN

' - - - - - - GND

HP/LINE - - - - - '

LHPIN----I
LLiNEIN - - - - I

>--+--t------

LOUT+

>--+-------

LOUT-

LIN - - - - - - - - - - - - - 1 -..

~TEXAS

3-566

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0212
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

AVAILABLE OPTIONS
PACKAGED DEVICE
TA

TSSOpt
(PWP)

-40°C to 85°C

TPA0212PWP

t The PWP package is available taped and
reeled. To order a taped and reeled part, add
the suffix R to the part number (e.g.,
TPA0212PWPR).

Terminal Functions
TERMINAL
NAME

NO.

DESCRIPTION

1/0

BYPASS

11

GAINO

2

I

Bit 0 of gain control

GAIN1

3

I

Bit 1 of gain control

GNO

Tap to voltage divider for internal mid-supply bias generator

1,12,
13,24

Ground connection for circuitry. Connected to the thermal pad.

LHPIN

6

I

LIN

10

I

Left channel headphone input, selected when SE/BTL is held high
Common left input for fully differential input. AC ground for single-ended inputs.

LLiNEIN

5

I

Left channel line input, selected when SE/BTL is held low

LOUT+

4
9

0
0

Left channel positive output in BTL mode and positive output in SE mode

LOUT-

Left channel negative output in BTL mode and high-impedance in SE mode

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a> 1-V (peak-to-peak) square wave is input
to PC-BEEP or PCB ENABLE is high.

HP/LINE

17

I

HP/LINE is the input MUX control input. When the HP/LINE terminal is held high, the headphone inputs
(LHPIN or RHPIN [6, 20]) are active. When the HP/LINE terminal is held low, the line BTL inputs (LLINEIN
or RLiNEIN [5, 23]) are active.

PVOO

7, 18

I

Power supply for output stage

RHPIN

20

I

RIN

8

·1

RLiNEIN

23

I

Right channel line input, selected when SE/BTL is held low

ROUT+

21

0

Right channel positive output in BTL mode and positive output in SE mode

ROUT-

16

0

Right channel negative output in BTL mode and high-impedance in SE mode

SHUTDOWN

22

I

Places entire IC in shutdown mode when held low, except PC-BEEP remains active

SE/BTL

15

I

Hold SE/BTL low for BTL mode and hold high for SE mode.

VOO

19

I

Analog VOO input supply. This terminal needs to be isolated from PVOO to achieve highest performance.

Right channel headphone input, selected when SEIBTL is held high
Common right input for fully differential input. AC ground for single-ended inputs.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-567

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284- NOVEMBER 1999

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, VOO ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to VOO +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-airtemperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, T J .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ........................... . . .. 260°C

t Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those Indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

DERATING FACTOR

PWP

21.8mWI"C

1.7W

1.4W

:): Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the Information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voRage, VOO
, High-level input voltage, VIH

MIN

MAX

4.5

5.5

SElBTL, HPILINE

4

SHUTOOWN

2

SElBTL, HPILINE

Low-level input voltage, VIL

0.8

Operating free-air temperature, TA

-40

V
V

3

SHUTOOWN

UNIT

85

V

°C

electrical characteristics at specified free-air temperature, Voo = 5 V, TA = 25°C (unless otherwise
.
noted)
PARAMETER

TEST CONDITIONS

MIN

TYP

MAX
25

UNIT

Output offset voitage (measured differentially)

VI = 0, Ay=-2VN

Power supply rejection ratio

VOO=4 Vt05 V

IIIHI

High-level input current

VOO = 5.5 V,
VI=VOO

900

nA

IIILI

Low-level input current

VOO=5.5V,
VI=OV

900

nA

100

Supply current

IVool
PSRR

BTL mode

6

8

SEmode

3

4

150

300

IOO(SO) Supply current, shutdown mode

~1ExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

mV
dB

77

mA

IJA

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284 - NOVEMBER 1999

operating characteristics, VDD

=5 V, TA =25°C, RL =8 Q, Gain =-2 VN, BTL mode

PARAMETER

TEST CONDITIONS
THO=l%,
RL=40

f= 1 kHz,
f=20Hzt015kHz

Po

Output power

THO+N

Total harmonic distortion plus noise

PO=1 W,

BOM

Maximum output power bandwidth

THO =5%

Supply ripple rejection ratio

f= 1 kHz,
CB = 0.47 I1F

SNR

IBTLmode

Signal-to-noise retio

Vn

Noise output voltage

ZI

Input impedance

CB=0.47 I1F,
f = 20 Hz to 20 kHz

MIN

TYP

MAX

UNIT

1.9

W

0.75%
>15

kHz

68

dB

105

dB

LBTLmode

16

I SE mode

30

I1V RMS

See Table 1

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power

1,4-7,10-13,
16-19,21

vs Frequency

2, 3, 8, 9, 14,
15,20,22

THO+N

Total harmonic distortion plus noise

Vn

Output noise voltage

vs Bandwidth

24

Supply ripple rejection retio

vs Frequency

25,26

Crosstalk

vs Frequency

27-29

Shutdown attenuation

vs Frequency

30

Signal-to-noise ratio

vs Frequency

vs Output voltage

SNR

Closed loop respone
Po
Po

Output power
Power dissipation

23

31
32-35

vs Load resistance

38,37

vs Output power

38,39

vs Ambient tempereture

40

~TEXAS

INSTRUMENTS
POST OFACE BOX 655303 • DALlAS, TEXAS 75265

3-669

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284-NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

10%

10%
AV=-2VN
f=1kHz
BTL
.1.

.~0
Z

+

I

c

I

II

0

'E

RL=40!

1%

i

~
r--

.~

0.1%

I==::

===

~-

'---

J

~0

:ii
Q

AV=-24 VN V
/1-'

1%

u

AV= 12VN

C
0

/

i

P

J

~

+

c

RL=30

I

!

z

I

I

I

§
as
:z:

I

RL=80

0

PO=1.75W
RL=30
BTL

.~

:z:

!

{!.

0.1

Av=-2VN

1./1"'~ V~

",'

~

/

{!.

I

I

Z

+

Z

Q

:z:

Q

~

+

:z:

I-

I-

0.01%
0.5 0.75 1 1.25 1.5 1.75 2

2.25 2.5 2.75

0.01%
20

3

100

Po - Output Power - W

AV =-6 VN

IIIIIII
1k

10k 20k

f - Frequency - Hz

Figure 1

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

RL=30

.~

Av=~2VN

BTL

z

--

+

I'

.,

c

~0
-;;
is

1%

r-..

f=15kHz

t-r-.

1

u

PO=1.0W

.....

-

ii 0.1%

~I

1

~llli

1k
f - Frequency - Hz

10k 20k

l-..J
RL=30
Av=-2VN
BTL

I-

0.01%
0.01

Figure 3

0.1
Po - Output Power - W

Figure 4

~TEXAS

3-570

11

:z:

.\

100

!-........' = 1 kHz

f=20Hz

Z
+
Q

PO=1.75W

0.01 %
20

r-

0

§
01
:z:

r7

PO=0.5W

''''

'2

1MV

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10%

10%

z~

f= 15kHz
~15kHZ

+

c

~

fl=U~1

I""-1"-0

J'""'o+.J.

,.......

E
!
~

1%

f=1kHz

~0

!

I ~

1%

I

-,.....
I I III r.J

1

"

f=1201~ I

...... r--

0.1%

,...

I-'

~~2~~1

0.1%

I

Z
+
C

r- RL=3Q

j!:

RL=3Q
- AV=-12VN
BTL

f-- AV=~VN
BTL

0.01%
0.01

0.01%
0.01

0.1
Po - Output Power - W

0.1
Po - Output Power - W

Figure 6

FigureS
TOTAL HARMONIC DISTORTION PLUS NOISE

vs

OUTPUT POWER

FREQUENCY
10%

f = 15 kHz

z~

~

I
I!

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10%

PO=1.75W
RL=3Q
BTL

I

+
c

~

"'"

1%

t""----,

~

1=1 kHz

rr--

0

0

t"--....

+

C

::c

AV= 12VN

f=20Hz

i"'r0.1

0.1%

I
Z

I-

AV=-24VN

1%

"2

~

10

OLr--

./

\

V

V

",II

V

.......

Av=-2VN

V

v

I

- RL=3Q
- AV=-24VN
BTL

0.01%
0.01

/~

0.1
Po - Output Power - W

0.01%
20

jVliUill
100

1k

10k 20k

f - Frequency - Hz

FigureS

Figure 7

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-571

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

100/0

..

10%

RL=40
AV= 2VN
BTL

.!!!

z0

RL=40
Av=-2VN
BTL

.~0

z

+

........

+

c

c

~

~

1%

ic

S

0

..

.2
c

:I:

0.1'%

!

..

E

~ '\

PO= 0.25 W

~

1

+

i==

0.01%
20

!

~

I'i+

0.1%

1

+

C
:I:
I-

100

10k 20k

1k

0.01%
0.01

0.1
Po- Output Power - W

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

OUTPUT POWER

OUTPUT POWER
10%

.~
z0

r--

+

c
0

r-f =15kHz

~ ~ 1151~~~ I

...,I

IJ

1%

0

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
10%

'Iii
is
()
';:

10

Figure 10

Figure 9

i:

J

f=20Hz

Z

.L. ~

C
:I:
I-

f= 1 kHz

:I:

PO=1.0W

Z

r-

0

. . . :%1-"

E

1

f=15kHz

is

PO=1.5W

.2
c

r-.

1%

1%

rf=1 kHz

~

0

i

f = 1 kHz

II I

I'r-

:I:

!

.."

,...

IIII

1'..... f ~ dO'~~ 1

i

0.1%

~

........

"I""tt+l J

f"'-."

0.1%

f=20Hz

1

Z

+

RL=40
AV =-6 VN
BTL

C
:I:
I-

0.01%
0.01

RL=40
AV=-12VN
BTL

I I IIII

~ ~
0.01%
0.01

0.1
Po - Output Power - W

Figure 11

0.1
Po - Output Power - W

Figure 12

~TEXAS

3-572

1111111

INSTRUMENTS
POST OFFICE BOX 555303 • DALLAS. TEXAS 75255

10

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

vs

OUTPUT POWER

FREQUENCY
10%

10%

..
II

RL=8Q
AV=-2VN
BTL

f = 15 kHz

'0

z

+

c
0

-

rNoL

1%

I

JJ~

1-1"-

'f

i..

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

fJ

c0
E
01

"

:c

]j 0.1%
~
I

~~~II

1%

f=20Hz

~

0.1%

~~

Po = 0.25 W

lie

PO= 1.0 W

Z

+

RL=4Q
AV=-24VN
BTL

C

:c

I-

"" ~

I I 1111111
0.1

0.01%
0.01

:J- '/
/'

0.01%
20

, 1

r-r--

---

100

Po - Output Power - W

,I

PO=0.5W
1k

10k 20k

f - Frequency - Hz

Figure 14

Figure 13
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

PO=1 W
RL=8Q
BTL

I'"""-..

~

AV=-24VN ""

z~

r=

RL=8Q

~

BTL

f- AV=-2VN

+
c

~

1%

~

r-. t---r-.

f = 15 kHz

..

is

/

L

AV =-12 VN

",I'-

1"- j/

v

C
0

"" Av=-2V/'V

~ III
V

Av=-eVN

....
0.01%
20

1-

:c

I

;§
I

-"""" &

Z

+

c

IL

i!:
1k

t--!.= 1 kHz

'OJ 0.1%

/'

100

E
01

10k 20k

f = 20 Hz

11111
0.01%

0.D1

f - Frequency - Hz

IIIII
0.1
Po - Output Power - W

10

Figure 16

Figure 15

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-573

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10%

10%

~

RL=80
~ AV=-6VN
~ i"--BTL
I"'"

Iz

+
c

_~=15kHZ

I

r-- '=15kHz

0

;:

i""-r--

1%

i

~0

--

:z::

l"- i""-

..E

ii

0.1%

~

1%
'=1 kHz
--,........

......

r-!=1 kHz

n

I I ll"t'H

I ,~Jol~1

0.1%

'=20Hz

I

Z

0
:z::

- RL=80
_ AV=-12VN
BTL

...

0.01%
0.01

0.01%
0.01

0.1
Po - Output Power - W

'"

0.1
Po - Output Power - W

Figure 17

10

Figure 18

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

va

vs

OUTPUT POWER

FREQUENCY

10%

10%

~z

RL=320
Av=-1 VN

I

1= 15 kHz '

j'"oooo

+

SE

+

I
I

c

~

1%

i

r-.
1=1 kHz

~

I

,~Jol~

ii

"'f't...

0.1%

~

ii

1%
pO=25mw~

P'"
0.1%

7

I

Z
+
CI

i

r- RL=80

...
:z::

r-

0.01%
0.01

Po=50mW

AV=-24 VN
BTL
0.1
Po - Output Power - W

10

~

0.01%
20

-.
100

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

\I

jOrlinITi -

1k
, - Frequency - Hz

Figure 20

Figure 19

3-574

~

10k 20k

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284-NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

100/0

3:
z

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY
100/0

F AV=-l
RL=320
VN
tt-

+

RL=10kO
Av=-l VN

I

t-

'0

SE

SE

+

c

g

~

!

is
£
c

"'-

0

..
III

:E:
'iii

0.1%

~I

1%

J

1%

fJl

f

Z
+
Q

£

g

f = 15 kHz
~

I

JH~

IIf 0.01%
.:!i

j!:

I

0.01%
0.01

0.001%
0.1

100

20

Po - Output Power -

w

OUTPUT NOISE VOLTAGE

vs

vs

OUTPUT VOLTAGE

BANDWIDTH

10%

100

RL = 10 kG
AV=-lVIV

VOO=5'V "'
90 I-R =40

SE

>:I.

+

I

1%

!

80

I
II
DI

70

~

80

!

is

.2
c 0.1%

Frequency - Hz

Figure 22

TOTAL HARMONIC DISTORTION PLUS NOISE

Z

10k 20k

lk
f-

Figure 21

.~

.-

VO=l VRMS

z

f=20Hz

I T1""H-I

j!:

0.1%

"'"

y

f=15kH

"""""""

z+

0

50

'S

40

z

f=2OHz

:!

IIf 0.01%

AV=-24VN

II

.!!!

!

~=~

I

AJI~ -12Iv~ 1\ ~
III jll'

30

AV=-tJVN .",.

c

>

Q

j!:

20

~i-"

10

0.001%
0.1

3

o

Vo - Output Voltage - VRMS

V

I"""

AV =-2 VN

~

10

~N.

~ i"'" ~

/

100

1k

10k

BW - Bandwidth - Hz

Figure 23

Figure 24

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS. TEXAS 75265

3-575

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284-NOVEMBER 1999

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO

SUPPLY RIPPLE REJECTION RATIO

vs

vs

FREQUENCY

FREQUENCY

o

rg

o

RL=SO
CB = 0.47 IJl'
BTL

-20

-20

ID
'0

I

jc

I

o

~

~O

i

i

, ,I ,111111
""'

t'--

..........

V

i-'"

IV

-80

J

-100

100

1k
f - Frequency - Hz

-120
20

10k 20k

1k

100

10k 20k

f - Frequency - Hz

Figure. 25

Figure 26

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

0

0
PO=1W
RL=SO
Ay =-2 VN
BTL

-20

-40

ID

AV=-1 VN ...

-60

8:
ii:

-100

-20

1"'r-.

.!!!

AV=-2VN

-rn

-120
20

-40

- I

AV =-24 VN

-60

r---r-.

I

I

-40

RL=320
CB=0.47 I1F
SE

ID

'0

PO=1W
RL=SO
Av =-24 VN
BTL

-40

'0

I

I

i

.oc
iii

-60

ie

(J

"""'-

-80
LEFT TO RIGHT
-100

100

1k
f - Frequency - Hz

-80

V

~

V

RI~~~OI~~~
-100

10k 20k

-120
20

111111
100

1k
f - Frequency - Hz

Figure 28

Figure 27

~TEXAS

INSTRUMENTS
3--576

I--

LEFT TO RIGHT

(J

I LUI
~
RI~tfr~O~

-120
20

..... ~

-60

POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

10k 20k

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284-NOVEMBER 1999

TYPICAL CHARACTERISTICS
CROSSTALK

SHUTDOWN ATTENUATION

vs

vs

FREQUENCY

FREQUENCY

0

0
VO=1 VRMS
RL = 10 k.Q
Av=-1 VN
SE

-20

III

"
I"'e"

VI=1 VRMS

I

-20

-40

III

-40

i

-60

"cI

I

-60

::I

RL=32n,SE

C

()

!

LEFT TO RIGHT,L

-80

...... .... ""

r::=::::::

-

-100

""

V

-80

~i'
~

-100

RIGHT TO LEFT

-120
20

II ilill
RL=10kn,SE

I I

100

1k

n-!!LmiY

1111111

-120

10k 20k

20

100

f - Frequency - Hz

1k

10k 20k

f - Frequency - Hz

Figure 29

Figure 30
SIGNAL-TQ-NOISE RATIO

vs
FREQUENCY
140
PO=1W
RL=SQ
BTL

130
III

"
I

ia:
~=

Iz
ic

120
110

~
a:
z

80

AV=-6VN

AV =-2 VN

I-

100 r-90

I

II
!!II::

j

II11

~

~

AV=-24VN

\

r--

~

~

AV=-12VN -

UJ

70

60
20

100

1k

10k 20k

f - Frequency - Hz

Figure 31

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-577

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE

10

Ill!llil

7.5

C"'I

5

III
"0
I

c

iii

2.5

r--

90°

Phase

0

1\

CI

-2.5
RL=8Q
AV =-2 VN
BTL

-5

.. I-.lUIWL

-7.5

-10
10

-900

U

tJ I I100I I

1k

10k

-180"

100k

1M

f - Frequency - Hz

Figure 32

CLOSED LOOP RESPONSE

30
25

20
Gain
III
"0

~

~

15

10

/

-

IJ~~~

r\

~

5

o
-5

r\

RL=8Q
AV =-6 VN
BTL
1111

-10 111111111
10
100

I

1k

10k

100k

f - Frequency - Hz

Figure 33

~TEXAS

3-578

-900

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

1M

-180°

TPA0212
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE
180'

30
111111

25

Gain

90'

20

15

I~~
II

i\

Phase

I'-

r--.

r\

5

o

"\

RL=8Q
AV=-12V/v
BTL
LUI LUll I JJ

IIIIIII II

-10
10

100

lk

10k

-90'

-180'

lOOk

1M

f - Frequency - Hz

Figure 34

CLOSED LOOP RESPONSE
30

180'

Gain

25
20

m

i' ..

'C

I

~

90'

\

Ijr-15
10

1\

Phase

r-..

5

o

"\

RL=8Q
AV =-24 V/V
BTL

-5

11111

-10
10

11111111
100

I

II

lk

10k

lOOk

-180'
1M

f - Frequency - Hz

Figure 35

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-579

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284-NOVEMBER 1999

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

va

va

LOAD RESISTANCE

LOAD RESISTANCE

3.5
3 J
~

2.5

'S
I:L
'S

1250

,

2

~I

.J

10%THD+N

0
D-

750

~

~

0

I

,p
0.5

o

0

1%TH~~
I
a

1

1000

'S

1.5

o

Av=-1 VN
SE

l\

I

J

1500

AV =-2 VN
BTL

R
i""ooo

~ 10%THD+N

500

\~

250
1%

16
24
32
40
48
RL - Load Resistance - 0

56

o

64

I
o

TH';:~ '"
I

a

16
24
32
40
48
RL - Load Resistance - 0

Figure 36

POWER DISSIPATION

va

va

OUTPUT POWER

OUTPUT POWER

1.a

~

1.4

I

I

c
0

i

1.2

30-

--

~

//V
J 1/
I V
e

0.4

-

~

L

0.6

D-

0.4

r/

I

0

J
I

D-

e

D-

1=1 kHz
BTL
Each Channel
1.5

2

0.2
0.15
0.1
0.05

2.5

I

0.25

I.

a~

0.5

./

0.3

c

40

0.2

o
o

0.35
~

o.a

DI

~

-"""i'oo.

V

......

1

'LV ~

o
o

I"-

f= 1 kHz

320

~

~

SE

Each Channel
~

U

M

U

Po - Output Power - W

Figure 39

~lEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. ~s 75265

K

."ao

I

Figure 38

40-

.............

~

Po - Output Power - W

3-Oao

64

Figure 37

POWER DISSIPATION

1.6

56

~

~

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284-NOVEMBER 1999

TYPICAL CHARACTERISTICS
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7

~JA11= 45.~oC~

\

ElJA4

6

\

~
I

c
.S!
'Oa.J

5

Ui

4

Iii
~

3

is

Q

"- 1\

\,
1\
~~
........

2

o

1\

""'" '- ~~
1""-

ElJA1,2

I

a.

.......

jJA3,

a.

ElJA2 = 45.2°CIW _
ElJA3 = 3l.2°CIW
ElJA4 = l8.6°CIW

~

~40

0

~

~

~

~

"

l00l~l~l~

TA - Ambient Temperature - °C

Figure 40

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STEREO 2·WAUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284 - NOVEMBER 1999

THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 41)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Side View (a)

End View (b)

Bottom View (e)

Figure 41. Views of Thermally Enhanced PWP Package

APPLICATION INFORMATION
selection of components
Figure 42 and Figure 43 are schematic diagrams of typical notebook computer application circuits.

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STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

Right CIRHP
Head- 0.47 I1F
phone
Input
Signal
20

-1

CIRLINE
Right 0.47 I1F
Line
Input
Signal

23

RHPfN
RLiNEIN

R
MUX

-1

8
CRIN
0.4711F

ROUT+

21

ROUT-

16

RIN

T

-=

PC BEEP
14
Input
Signal epCB
0.47 11F

-J

PC-BEEP
PCBeep

100kn
2

GAINO

3

PVDD
Depop
Circuitry

Left CILHP
Head- 0.47 I1F~'-t-~==-_.....I
phone
Input
Signal

-7

Power
Management

18

VDD

19

BYPASS
SHUTDOWN

11

GNO

See Note A
VDD
CSR

1;:' 0.111F

VOO

T

-=

22

P
-=

CSR
0.111F

CBYP

1;:' 0.4711F

To
SystemControl

LOUT+

4

LOUT-

9

1 kn

1,12,
13,24

-=

-=

COUTL
330I1F

LIN

100kn
NOTE A.

A 0.1 I1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 I1F or greater should be placed near the audio power amplifier.

Figure 42. Typical TPA0212 Application Circuit USing Single-Ended Inputs and Input MUX

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STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION
N/C
CIRINRight 0.47 ~F
Negative _\
23
DifferentiaO 1--=+-'=='-'--1
Input
Signal

CIRIN+
Right 0.47 ~F
Positive --'l
8

ROUT+

21

RIN

DifferentiaI71-~1--'-''''-'------+''
Input
Signal

COUTR

PC BEEP
Input
14
Signal CpCB

---j

330~F

PC-BEEP

PC-

ROUT-

16

VDD

Beep

OA7~F

-=-

1kf.!

100kf.!
2
3

GAl NO
GAIN1

15

SElBTL

Gainl
MUX
Control

NlC
6

LHPIN

5

LLINEIN

CILINLeft
Negative 0.47 ~F
Differential ~
Input
Signal
CILlN+
Left 0.47~F
Positive ~
10
Differential
Input
Signal

18

VDD

19

BYPASS
SHUTDOWN

11

Depop
Circuitry
Power
Management

HP/LINE

PVDD

GND
L
MUX
LOUT+

See Note A
VDD
CSR
1='0.1 ~F
VDD

T

CSR
0.1~F

22

CBYP
To 1=' 0.47 ~F
System
Control
1,12,
4
13,24

1 kf.!

COUTL
330~F

LIN

LOUT-

9

100kf.!
NOTE A.

A 0.1 ~F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 ~F or greater should be placed near the audio power amplifier.

Figure 43. Typical TPA0212 Application Circuit Using Differential Inputs

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STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION
gain setting via GAl NO and GAIN1 inputs
The gain of the TPA0212 is set by two input terminals, GAINO and GAIN1.
Table 1. Gain Settings
GAINO

GAIN1

SE/BTL

0

0

0

0

1

0

1
1

0

0

1

0

X

X

1

Av
-2VN

-6VN
-12VN
-24VN
-1VN

The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, Z" to be dependant on the gain setting. The actual gain settings are controlled
by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input
impedance will shift by 30% due to shifts in the actual resistance of the input impedance.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 10 kQ, which is the absolute minimum input impedance of the TPA0212. At the higher gain
settings, the input impedance could increase as high as 115 kn.
input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r------------

I
I
I

ZF

C
ZI
IN
Input --------'If------.-.:.:.:..+--A,/\Iv-*--l

Signal ----------;

R

The typical input impedance at each gain setting is given. in the table below:

Av

ZI

-24VN
-12VN
-6VN
-2VN

14 kO
26kO
45.5 kO
91 kn

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STEREO 2·W AUDIO POWER AMPLIFIER
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SLOS284- NOVEMBER 1999

APPLICATION INFORMATION
The -3 dB frequency can be calculated using equation 1:

f

-

-3

1

dB - 2It C(R II R,)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addition, the order of the filter could be increased.

input capacitor, C,
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, C, and the input impedance of the amplifier, Z" form a
high-pass filter with the corner frequency determined in equation 2.

(2)
fC(hlghpaSS) =

23ti, C,

The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where Z, is 710 k.Q and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
C -

1

, - 23tZ, fc

(3)

In this example, C, is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 1lF. A further
consideration for this capacitor is the leakage path from the input source through the input network (C,) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc 'evel. Note that it is important to confirm the capacitor polarity in the application.

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STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION
power supply decoupling, Cs
The TPA0212 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THD) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 ~F placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 ~F or greater placed near
the audio power amplifier is recommended.
midrail bypass capacitor, CBYP
The mid rail bypass capacitor, CBYP, is the most critical capacitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CBYP, values of 0.47 ~F to 1 ~F ceramic or tantalum low-ESR capacitors are recommended
for the best THD and noise performance.

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter govemed by equation 4.

(4)

fC(hlgh)

The main disadvantage, from a performance standpOint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 ~F is chosen and loads vary from 3 n,
4 n, 8 n, 32 n, 10 kn, to 47 kn. Table 2 summarizes the frequency response characteristics of each
configuration.

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STERE02-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL

SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode

Cc

Lowest Frequency

3Q

33Ol1F

161 Hz

4Q

33011F

120 Hz

8Q

330l1F

60Hz

32Q

33Ol1F

15Hz

10,000Q

330l1F

0.05 Hz

47,000Q

33Ol1F

0.01 Hz

RL

As Table 2 indicates, most of the bass response is attenuated into a 4-n load, an 8-n load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

bridged-tied load versus single-ended mode
Figure 44 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0212 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but initially consider power to the load. The differential drive to the speaker
means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the
voltage swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power
equation, where voltage is squared, yields 4x the output power from the same supply rail and load impedance
(see equation 5).

v

_ VO(PP)

(rms) -

(5)

2./2

2
V(rms)

Power = - RL

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STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION

voo

J'
RL

~

J'!
rv ~

VO(PP)

2x vO(PP)

-vO(PP)

Figure 44. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an 8-n speaker from a
singled-ended (SE, groul1d reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 45. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capaCitors can be quite large (approximately 331!F to 1000 I!F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
fc =

(6)

1

21tRL C c

For example, a 68-I!F capaCitor with an 8-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

Voo

~dB~----~~====

Figure 45. Single-Ended Configuration and Frequency Response

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STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S284 - NOVEMBER 1999

APPLICATION INFORMATION
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

single-ended operation
In SE mode (see Figure 44 and Figure 45), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier'S gain to 1 VN.

Input MUX operation
The input MUX allows two separate inputs to be applied to the amplifier. This allows the designer to choose
which input is active independent of the state of the SElBTL terminal. When the HPILINE terminal is held high,
the headphone inputs are active. When the HP/LINE terminal is held low, the line BTL inputs are active.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these ineffiCiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output pow~r. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The internal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 46).
100

,/

-~-

V(LRMS)

IOD{avg)

Figure 46. Voltage and Current Waveforms for BTL Amplifiers

Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

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STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION
Efficiency of a BTL amplifier =

p--'P =-

(7)

SUP

Where:
PL

=

T'
V rms 2

Vp

andV LRMS

= !2'

therefore, PL

=~

looavg

and

=

Vp 2
2RL

Jo" VRL sin(t) dt = ~
P

V
It
x RP [cos(t)] 0
L

=

2V
It :

L

Therefore,
_ 2 VOO Vp
PSUP It RL
substituting PL and Psup into equation 7,
Vp 2

Efficiency of a BTL amplifier

~

PL =Power devilered to load
Psup =Power drawn from power supply
VLRMS = RMS voltage on BTL load
RL = Load resistance
Vp =Peak voltage on BTL load
looavg =Average current drawn from
the power supply
Voo =Power supply voltage
llBTL =Efficiency of a BTL amplifier

ItVp

2Voo Vp = 4 Voo
It RL

Where:

Therefore,

l]BTL

(8)

Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.

Table 3. Efficiency Vs Output Power in 5-V 8-0 BTL Systems
Output Power

Efficiency
(%)

Peak Voltage
(V)

Internal Dissipation

(W)

0.25

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.47t

0.53

(W)

t High peak voltages cause the THO to Increase.
A final pOint to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

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STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL

SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION

crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the TPA0212 data sheet, one can see
that when the TPA0212 is operating from a 5-V supply into a 3-Q speaker 4-W peaks are available. Converting
watts to dB:

P

P dB = 10Log~ = 10Log 4 W = 6 dB
P ref
1 W

(9)

Subtracting the headroom restriction to obtain the average listening level without distortion yields:

6 dB - 15 dB = -9 dB (15 dB crest factor)
6 dB - 12 dB = --6 dB (12 dB crest factor)
6 dB - 9 dB = -3 dB (9 dB crest factor)
6 dB - 6 dB =0 dB (6 dB crest factor)
6 dB - 3 dB =3 dB (3 dB crest factor)
Converting dB back into watts:
1QPdB/10

x

P

ref

(10)

63 mW (18 dB crest factor)
125 mW (15 dB crest factor)
= 250 mW (9 dB crest factor)
= 500 mW (6 dB crest factor)
= 1000 mW (3 dB crest factor)
= 2000 mW (15 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for a 5-V, 3-Q system, the internal dissipation in the TPA0212 and
maximum ambient temperatures is shown in Table 4.

Table 4. TPA0212 Power Rating, 5-V, 3-Q, Stereo
PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

4

2W(3dB)

1.7

-3°C

4

1000 mW (6 dB)

1.6

6°C

4

500mW(9dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

78°C

4

63mW(18dB)

0.6

96°C

(W)

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STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION

crest factor and thermal considerations (continued)
Table 5. TPA0212 Power Rating, 5-V,

S-n. Stereo

PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

2.5W

1250 mW (3 dB crest factor)

0.55

100°C

2.5W

1000 mW (4 dB crest factor)

0.62

94°C

2.5W

500 mW (7 dB crest factor)

0.59

97°C

2.5W

250 mW (10 dB crest factor)

0.53

102°C

The maximum dissipated power, PDmax, is reached at a much lower output power level for an 8-Q load than for
a 3-Q load. As a result, this simple formula for calculating PDmax may be used for an 8-Q application:

P

2Vfm

Dmax

(11 )

=--

:n;2R

L

However, in the case of a 3-Q load, the PDmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the PDmax formula
for a 3-Q load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to 9JA:

e

JA

=

1
Derating Factor

=

_1_
0.022

=

450C/W

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given 9JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0212 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
T A Max = T J Max -

=

e JA

(13)

PD

150 - 45(0.6 x 2)

=

96°C (15 dB crest factor)

NOTE:
Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.

Tables 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0212 is deSigned with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-Q speakers dramatically increases the thermal performance by increasing amplifier
efficiency.

-!II

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STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS2ll4- NOVEMBER 1999

APPLICATION INFORMATION
SElBTL operation
The ability of the TPA0212 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
intemal stereo speakers are driven in BTL mode but extemal headphone or speakers must be accommodated.
Intemal to the TPA0212, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT- and ROUT- (terminals 9 and 16). When
SElBTLls held low, the amplifier is on and the TPA0212 is in the BTL mode. When SElBTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0212 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SElBTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 47.

20

RHPIN

23

RUNEIN

R

MUX
ROUT+

8

21

RIN

VDD
ROUT-

16

100kn
sEiafi:

15 100kn

~

n

~~
Figure 47. TPA0212 Resistor Divider Network Circuit

Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-kO./1-kn divider pulls the SElBTL input low. When a plug is inserted, the 1-kn
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the OUT-amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.

~TEXAS

INSTRl)MENTS
3-594

POST OFFICE BOX 665303 • DALLAS. TEXAS 75265

TPA0212
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS284 - NOVEMBER 1999

APPLICATION INFORMATION
PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is activated automatically. When the PC BEEP input is active,
both of the LlNEIN and HPIN inputs are deselected and both the left and right channels are driven in BTL mode
with the signal from PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is
independent of the volume setting. When the PC BEEP input is deselected, the amplifier will return to the
previous operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC
BEEP will take the device out of shutdown and output the PC BEEP signal, then return the amplifier to shutdown
mode.
The preferred input signal is a square wave or pulse train with an amplitude of 1 Vpp or greater. To be accurately
detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall times of less than 0.1 (.IS and a
minimum of 8 rising edges. When the signal is no longer detected, the amplifier will return to its previous
operating mode and volume setting.
If it is desired to ac-couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
equation 14:
C

>

PCB - 2/t

f pCB1 (100 kQ)

(14)

The PC BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.

shutdown modes
The TPA0212 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, 100 150 IJA. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.

=

Table 6. HP/LINE, SE/BTL, and Shutdown Functions
AMPLIFIER STATE

INPUTst

t

HP/LINE

SElBTL

SHUTDOWN

INPUT

OUTPUT

X

X

Low

X

Mute

Low

Low

High

Line

BTL

Low

High

High

Line

SE

High

Low

High

HP

BTL

High

High

High

HP

SE

Inputs should never be left unconnected.
X do not care

=

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-595

3-596

TPA0213
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS276B-

OGOPACKAGE
(TOP VIEW}

• Ideal for Notebook Computers, PDAs, and
Other Small Portable Audio Devices
• 2 W Into 4-0 From 5-V Supply
• 0.6 W Into 4-0 From 3-V Supply

MONO-IN
SHUTDOWN

• Stereo Head Phone Drive
• Separate Inputs for the Mono (BTL) Signal,
and Stereo (SE) Left/Right Signals

VDD

BYPASS
RIN

LO/MoLIN
GND
ST/MN
RO/MO+

• Wide Power Supply Compatibility 2.5 V to
5.5 V
• Low Supply Current
- 4.2 mA Typical at 5 V
- 3.6 mA Typical at 3 V
• Shutdown Control ••• 1 ~A Typical
• Shutdown Pin is TTL Compatible
• -40°C to 85°C Operating Temperature
Range
• Space-Saving, Thermally-Enhanced MSOP
Packaging

description
The TPA0213 is a 2-W mono bridge-tied-Ioad (BTL) amplifier designed to drive speakers with as low as 4-0
impedance. The amplifier can be reconfigured on-the-fly to drive two stereo single-ended (SE) signals into head
phones. This makes the device ideal for use in small notebook computers, PDAs, Digital Personal Audio
players, anyplace a mono speaker and stereo head phones are required. From a 5-V supply, the TPA0213 can
deliver 2·W of power into a 4-0 speaker.
The gain of the input stage is set by the user-selected input resistor and a 50-kQ internal feedback resistor
(Av = - RF/ RI). The power stage is internally configured with a gain of -1.25 VN in SE mode, and -2.5 VN in
BTL mode. Thus, the overall gain of the amplifier is 62.5 knt RI in SE mode and 125 knt RI in BTL mode.
The TPA0213 is available in the 10·pin thermally-enhanced MSOP package (DGO) and operates over an
ambient temperature range of -40°C to 85°C.

..

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

3-597

TPA0213
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS276B - JANUARY 2000 - REVISED MARCH 2000

4

1-:------------VDD

3

1

CI

Input

1~vV\r~

Right
Audio

CI

Input

II--J\I\I\,.--'

8

GND

50kQ
Mono
Audio

BvPAss--------l

VDD

1 kQ

VDD

BYPASS
50 kQ

1.25*R

1
1

50kQ
Stereo/Mono
Control

50kQ

STiMN

1

100kQ

7

1

1

50kQ

1

1.25*R

1

Left
Audio
Input

1

CI

1

1

1r-~RNI~_9-rIL_IN____~~

LO/MO-

1

1

1

1

1

1

BYPASS

1

From
System Control

1

21

1

1

SHUTDOWN

Shutdown
and Depop
Circuitry

1
1
1

1

L _________________________

~TEXAS

INSTRUMENTS
3-598

POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

Cc

1 10

1

~

1 kQ

TPA0213
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS276B - JANUARY 2000 - REVISED MARCH 2000

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

MSOpt
(DGQ)

-40°C to 85°C

TPA0213DGO

MSOP
SYMBOLIZATION
AEH

t The DGO package are available taped and reeled. To order a taped and reeled part, add the
suffix R to the part number (e.g., TPA0213DGOR).

Terminal Functions
TERMINAL
NAME

NO.

I/O

DESCRIPTION

MONO-IN

1

I

Mono input terminal

SHUTDOWN

2

I

SHUTDOWN places the entire device in shutdown mode when held low. TTL compatible input.

VDD

3

I

VDD is the supply voltage terminal.

BYPASS

4

I

BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected
to a 0.1-IlF to 1-IlF capacitor.
Right-channel input terminal

RIN

5

I

RO/MO+

6

0

ST/MN

7

I

Selects between stereo and mono mode. When held high, the amplifier is in SE stereo mode, while held
low, the amplifier is in BTL mono mode.

Left-channel input terminal

GND

8

LIN

9

I

LO/MO-

10

0

Right-output in SE mode and mono positive output in BTL mode

Ground terminal

Left-output in SE mode and mono negative output in BTL mode.

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maXimum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maxi mum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
DGO

DERATING FACTOR
2.14m1

17.1 mWrC

TA
1.37W

= 85°C

1.11 W

11 Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMAOO2), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-599

TPA0213
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
Sl0S276B - JANUARY 2000 - REVISED MARCH 2000 _

recommended operating conditions
Supply voltage, VOO
STIMN

High-level Input voltage, VIH

STIMN

MAX

2.5

5.5

lVOO=3V

2.7

IVOO=5V

4.5

SHUTDOWN
Low-level Input voltage, VIL

MIN

UNIT
V
V

2

I VOO=3V

1.65

I VOO=5V

2.75

SHUTDOWN

V

0.8

Operating free-air temperature, TA

-40

·C

85

electrical characteristics at specHled free-air temperature, VDD =3 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

IVool

Output offset voltage (measured differentially)

VIO=O,

Galn=8dB

PSRR

Power supply rejection ratio

VOO=2.9Vto3.1 V,

BTL mode

IIIHI

High-level input current

VOO=3.3V,

VI=VOO

VOO = 3.3 V,

VI=O

MIN

TVP

MAX

UNIT

30

mV

1
1

I1A
I1A

65

dB

IIILI

Low-level Input current

Zi

Input Impedance

50

100

Supply current

3.6

5.5

mA

IOO/SO)

Supply current, shutdown mode

1

10

I1A

operating characteristics, VDD

=3 V, TA =25°C, RL =4 Q, f =1 kHz (unless otherwise noted)

PARAMETER

TEST CONDITIONS
THO = 1%,

BTL mode

THO=0.1%,

SEmode,

Po

Output power, see Note 1

THO+N

Total hannonic distortion plus noise

Po=500mW,

f=20 Hz to 20 kHz

BOM

Maximum output power bandwidth

Gain=8dB,

THO=2%

Supple ripple rejection ratio

Vn

Noise output voltage

f= 1 kHz,

CB=0.47 I1F,

CB = 0.47 I1F

f=20 Hz to 20 kHz

MIN
RL=320

33

MAX

UNIT
mW

0.2%
20
BTL mode

52

SEmode

62

BTL mode

42

SEmode

21

~1ExAs

INSTRUMENTS

TVP

860

NOTE 1: Output power Is measured at the output tenninals of the device at f = 1 kHz.

3-600

kn

POST OFRCE sox 655303 • DALLAS, TEXAS 75265

kHz
dB

I1V RMS

TPA0213
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS276B - JANUARY 2000 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, VDD
noted)
PARAMETER

=5 V, TA =25°C (unless otherwise

TEST CONDITIONS

MIN

TYP

MAX

UNIT

IVool

Output offset voltage (measured differentially)

VIO=O,

Gain=8dB

PSRR

Power supply rejection ratio

VOO =4.9Vt05.1 V,

BTL mode

IIIHI

High-level input current

VOO=5.5V,

VI=VOO

1

IlA

IIILI

Low-level input current

VOO=5.5V,

VI=O

1

I!A

ZI

Input impedance

50

100

Supply current

4.2

6.3

rnA

IOO(SO)

Supply current, shutdown mode

1

10

I!A

operating characteristics, VDD

62

mV
dB

kQ

=5 V, TA =25°C, RL =4 n

PARAMETER

TEST CONDITIONS

THO = 0.3%,

BTL mode

THO=O.l%,

SEmode,

Po

Output power, see Note 1

THO+N

Total harmonic distortion plus
noise

PO= 1.5W,

f = 20 Hz to 20 kHz

BOM

Maximum output power bandwidth

Gain=6dB,

THO=2%

Vn

30

Supple ripple rejection ratio

f= 1 kHz,

CB = 0.47 IlF

Noise output voltage

CB = 0.47 IlF,

f = 20 Hz to 20 kHz

MIN

RL=32Q

TYP

MAX

UNIT

2

W

90

mW

0.2%
20
BTL mode

52

SEmode

62

BTL mode

42

SEmode

21

kHz
dB

IlV RMS

NOTE 1: Output power is measured at the output terminals of the device at f = 1 kHz.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-601

TPA0213
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S276B - JANUARY 2000 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION + NOISE
va
OUTPUT POWER

TOTAL HARMONIC DISTORTION + NOISE
va
FREQUENCY

10

I

I
~

r- YDD=;;V

i= Mono/BTL

Iz

r- Mono/BTL
r-f=1 kHz
r- Galn=8dB

+
c

J

1

I- VDD=3Y

r- RL=8C
r- Po =250mW

+
c
0

i!

.....

R =

......

~
.10 "=

........
C

=

O.1

I
I

tialn =

oS!
c

n

!'--

.....

~

~

0

llln= dB
0.01

]

{!.

I

I

Z

Z

~

~
:c

j!:

1-,

.01
0.001

0.01
0.1
1
Po - Output Power - W

0.001
10

10

1k

100

10k 20k

f - Frequency - Hz

Figure 1

Figure 2

TOTAL HARMONIC DISTORTION + NOISE
va
OUTPUT POWER

TOTAL HARMONIC DISTORTION + NOISE
va
FREQUENCY

10
.---:VDD=3Y

~

0

i!
0

t= = 21 k

oS!

~I
z

~

~

II

1-00.

r-. ~

0.1

=1

0.1

]
{!.

:nz

I'~

l/

0.01

I

-

j!:
0.01
0.001

I~=:21
I-

L= OkC

Z

I ~q ~11IrR"'S

+

-I

CI

:c

I

I-

0.01

2

0.1

0.001
10

Po - output Power - W

11I1111
100

1k

f - Frequency - Hz

Figure 4

Figure 3

~TEXAS

INSTRUMENTS
3-602

V- ....

~,

~
:c

II

L=32C

P =25mW

oS!
c

~

c0

t - StereolSE
t - Gain = 1.9 dB

••zc+

Gain=8dB
i""'~r-I

+

c

I
I

~VDD=3V

~Mono/BTL
I-- RL=8n

Iz

POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

10k 20k

TPA0213
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS276B - JANUARY 2000 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION + NOISE

TOTAL HARMONIC DISTORTION + NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10

10
~VOO=;iV

:

t- StereolSE
t- RL=320

'0
z
+

t-f=1 kHz
_ Gain =8dB

+

c

~

~0

i
~
.
::c

1i
Q

0

I§

li

~Mono/BTL

z

_ Gain = 1.9 dB

c

t- VOO=5V

.~

r-- f= 20 kHz

I

~

0.1

Ii

.....

::c

li

~I =

I

+
Q

Z

+

IL

::c

Q

::c

I-

f=20Hz
0.01
0.01

0.01
0.001

0.1

0.01

Po - Output Power - W

TOTAL HARMONIC DISTORTION + NOISE

vs

vs

FREQUENCY
10

-

VOO=l)V
Mono/BTL
-RL=80
_PO=1W

+

~

OUTPUT POWER

=

C

I
.
::c

0.1

li

0.01

110
z

h

~
.~

I'\.

c

0

~

I§

,...

~

r- VOO=5V
f:= Mono/BTL
r- RL=80

r---

+

c

0

:e

Gain = 8 dB

1"-1"f= O~z

I..

Ga n= Od

.S!

10

Figure 6

TOTAL HARMONIC DISTORTIG>N + NOISE

Z

0.1

Po - Output Power - W

Figure 5

110

.....

0.1

~

r------ f = 1 kHz

I-

=4'

I""'"

0

~

I
Z

'!!Ioo

.!:!
c

I

'2

E;;:-

0

Ii

an=8 B

::c

li

~

.....
1',..
""'"

r--. ,....-

0.1

~

I

I

Z

+

"""

,HZ

=

Z

z

+
Q

Q

::c

j!:

I-

0.001
10

100

1k

10k 20k

0.01
0.001

f - Frequency - Hz

0.01

0.1

2

Po - Output Power - W

Figure 7

FigureS

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3--603

TPA0213
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS2768 - JANUARY 2000 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION + NOISE

TOTAL HARMONIC DISTORTION + NOISE

vs

vs

FREQUENCY

OUTPUT POWER

=
r-

·z1

..

vDD=:JV
Stereo/SE
Gain = 1.9 dB

10
f- vDD=ll V

CD

~ Stereo/SE

'0

z

+

f-RL=32Q
Gain = 1.9dB

+

c

c

~0

0

'E0

0.1

'Iii

C
u

'c0

RL = 3 II
PO=75mW

,

1\'

III1\1

,

J:

~

0.01

S

{!.

~u

....

V

'c0

Ii

1/V""

S

Z

+

Q

...
J:

I

100

~ r-f=1

Z

+

...

1k

I

il=20 HZt

0.01
0.01

10k 20k

0.1

f - Frequency - Hz

Po - Output Power - W

Figure 9

Figure 10
POWER SUPPLY REJECTION RATIO
vs
FREQUENCY·

vs
. FREQUENCY
100

Mo~~/

Hz

I

J:

OUTPUT NOISE VOLTAGE

-.11

i"o ~

Q

IIIIII

0.001
10

............

0.1

{!.

R'L =10kQ
V
I Anl1 VRr~

I

=2 k:tz

J:

0

IIII
III

1111/

r- RL=8Q

Mono 11
RL=8Q

f- GaT

Gilnl = 8

r2°I~i

r-~eU/ IIII

/

V

RL=32Q
Gain=14dB

1/

-

iii -

.!te!e~s III _
RL=32Q
Gain = 1.9dB

"
I

0

-20

1i
a:
c
0

13CD

l

a
;

r-.... ~

ci

........ ~~

-40

........
-60

~i"o

=1

~~f'

~

I

=10~F

i--""

"' f""

::I

III

-60

" By ass = .5V

0

a..

a:
a:

Mono/BTL
Galn=8dB -

Vr--e---+-----

ROUT-

- - - - - - - - - + -....

PC-BEEP--i

GAINO
GAlN1

>--e-------

PCBeep

:=;:;::::::

Power
Management

SElBTL
L - -_ _ _

HP/UNE - - - - - '

PVDD
VDD
BYPASS
SHUTDOWN

GND

LHPIN---f
LUNEIN--~

UN

>--+---1-----

LOUT+

>-.._-----

LOUT-

- - - - - - - - - + -....

~TEXAS

INSTRUMENTS
POST OFRCE BOX 655303 • DAllAS. TEXAS 75265

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S285 - NOVEMBER 1999

AVAILABLE OPTIONS
PACKAGED DEVICE

t

TA

TSSOpt
(PWP)

-40°C to 85°C

TPA0222PWP

The PWP package IS available taped and
reeled. To order a taped and reeled part, add
the suffix R to the part number (e.g.,
TPA0222PWPR).

Terminal Functions
TERMINAL
NAME

NO.

DESCRIPTION

I/O

BYPASS

11

GAl NO

2

I

Bit 0 of gain control

GAINI

3

I

Bit 1 of gain control

GNO

Tap to voltage divider for internal mid-supply bias generator

1,12,
13,24

Ground connection for circuitry. Connected to the thermal pad

LHPIN

6

I

LIN

10

I

Left channel headphone input, selected when SElBTL is held high
Common left input for fully differential input. AC ground for single-ended inputs

LLiNEIN

5

I

Left channel line input, selected when SElBTL is held low

LOUT+

4

LOUT-

9

0
0

Left channel negative output in BTL mode and high-impedance in SE mode

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > I·V (peak-to-peak) square wave is input
to PC-BEEP or PCB ENABLE is high.

HP/LINE

17

I

HP/LINE is the input MUX control input. When the HP/LINE terminal Is held high, the headphone inputs
(LHPIN or RHPIN [6, 20]) are active. When the HP/LINE terminal is held low, the line BTL inputs (LLINEIN
or RLiNEIN [5, 23]) are active.

PVOO

7,18

I

Power supply for output stage

RHPIN

20

I

Right channel headphone input, selected when SElBTL is held high

Left channel positive output in BTL mode and positive output in SE mode

RIN

8

I

Common right input for fully differential input. AC ground for single-ended inputs

RLiNEIN

23

I

Right channel line input, selected when SElBTL is held low

ROUT+

21

0

Right channel positive output in BTL mode and positive output in SE mode

ROUT-

16

0

Right channel negative output in BTL mode and high-impedance in SE mode

SHUTDOWN

22

I

Places entire IC in shutdown mode when held low, except PC-BEEP remains active

SElBTL

15

I

Hold SElBTL low for BTL mode and hold high for SE mode.

VOO

19

I

Analog VOO input supply. This terminal needs to be isolated from PVOO to achieve highest performance.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-609

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ...................................................... ; ..... -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C

t

Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE

DERATING FACTOR
21.8mW/oC

2.7wl

PWP

1.7W

1.4W

:I: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
High-level input voltage, VIH

MIN

MAX

4.5

5.5

SElBTL, HPILINE

4

SHUTDOWN

2

SElBTL, HPILINE

LOW-level input voltage, VIL

0.8
-40

Operating free-air temperature, TA

V
V

3

SHUTDOWN

UNIT

85

V
°C

electrical characteristics at specified free-air temperature, VDD= 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

MIN

TYP

AV =-2 VN

MAX

Output ollset voltage (measured differentially)

VI=O,

PSRR

Power supply rejection ratio

VOO = 4.9 V to 5.1 V

IIIHI

High-level input current

VOO=5.5V,
VI=VOO

900

nA

IIILI

Low-level input current

VOO=5.5V,
VI=OV

900

nA

100

Supply current

IOO(SO)

Supply current, shutdown mode

77

BTL mode

18

SEmode

9
150

~lEXAS

INSTRUMENTS
3-610

25

UNIT

IVosl

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

mV
dB

rnA
300

!LA

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

operating characteristics, VDD

=5 V, TA =25°C, RL =8 n, Gain =-2 VIV, BTL mode

PARAMETER

TEST CONDITIONS

THD=l%,
RL=4n

f= 1 kHz,
f=20Hzto 15kHz

Po

Output power

THD+N

Total harmonic distortion plus noise

PO=l W,

BOM

Maximum output power bandwidth

THD=5%

Supply ripple rejection ratio

f= 1 kHz,
CB = 0.47!1F

SNR

I

BTL mode

Signal-to-noise ratio

Vn

Noise output voltage

Z,

Input impedance

CB=0.47!1F,
f = 20 Hz to 20 kHz

IBTL mode
I SE mode

MIN

TYP

MAX

UNIT

1.9

W

0.5%
>15

kHz

68

dB

105

dB

16
30

!1VRMS

See Table 1

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power

1,4-7,10-13,
16-19,21

vs Frequency

2,3,8,9,14,
15,20,22

THD+N

Total harmonic distortion plus noise

Vn

Output noise voltage

vs Bandwidth

24

Supply ripple rejection ratio

vs Frequency

25,26

Crosstalk

vs Frequency

27-29

Shutdown attenuation

vs Frequency

30

Signal-to-noise ratio

vs Bandwidth

vs Output voltage

SNR

Closed loop respone
Po
PD

Output power
Power dissipation

23

31
32-35

vs Load resistance

36,37

vs Output power

38,39

vs Ambient temperature

40

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

3-611

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS ANDMUX CONTROL
SL0S285-NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
vs·
OUTPUT POWER

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

10%

10%
AV=2VN
fe1 kHz
BTL

Iz

+

I

I:
0

;:

1%

j
0

I I

I

J

RL=40

I

r=
RL=eo
rI

.1

AV=-4.4VN

=

,--

I--

AV =-12 VN

~

0.1%

!z

'"

~

1,/

L /

I

I

.....

I

I

-'- RL=30

i!
0

PO='·75W
RL=30
BTL

I

0./

'"

~V

AV =-2 VN

V

I'll

I

0

111~V=1~~

~

0.01%
0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 2.75

0.01%
20

3

Po - Output Power - W

Figure 1

10%

RL=30
AV=-2VN
. BTL

J

~

+

J

!

+

V

'II.

PO=O.5W

V

~

j

1%

r- t""-

Ii

~~

~

t;'

r-0.1%

I

~

RL=30
AV=-4.VN
BTL

~

Ii 11111

1k

10k 20k

0.01%
0.01

,- Frequency - Hz

0.1
Po - Output Power - W

Figure 4

Figure 3

~1ExAs

H12

1=1 kHz

litt-.J..l

0

~\
PO='·75W

100

J

'=20Hz

Z

0.01 'II.
20

'=15kHz

~0

..... ~

PO='·0W
0.1IC!/

10k 20k

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
OUTPUT POWER

10%

I

II

1k
,- Frequency - Hz

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE
vs
FREQUENCY

I

1IIII
100

I

INSTRUMENTS
POST OFFICE BOX 655303 ;, DAUAS. TEXAS 75265

10

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTOFmON PLUS NOISE

TOTAL HARMONIC DISTOFmON PLUS NOISE

va

va

OUTPUT POWER

OUTPUT POWER

10%

10%

Iz

1= 15kHz

+

c

~

~

1%

I

-

iE

Iz

f= 15 kHz

I
I

V

1=1 kHz

r:---+-J. J.l

f=I20I~ r

S

~

I

0%

~

- RL=3n
- AV=-eYN
BTL

I-

0.01%
0.01

10

fl=ll

1%

I""'---.

f=20Hz

1--1-

0.1%

0.1

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

va

TOTAL HARMONIC DISTORnON PLUS NOISE

va

OUTPUT POWER

FREQUENCY

10%re!F1ll!m~

10%
.1

1/
1%

I

.!:!
c

h:1 kHz

.....

- ....

E

f=20Hz

%

~I

PO=1.5W
RL=4n
BTL

1= 15 kHz

~

S

10

Po - Output Power - W

FigureS

+
c

V

'- AV=-12VN
BTL

0.01%
0.01

Po - Output Power - W

Iz

JJT

II

r- RL=3n

i!:
0.1

~

+

J

1-00

r--.. r--..
0.1%

:!

I

0.1%

Z

0
i!:

- RL=3n
- Ay=-24YN
BTL

0.01%
0.01

0.1

10

Po - Output Power - W

f - Frequency - Hz

Figure 7

/

FigureS

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAU.AS, TEXAS 75285

3-e13

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

RL=40
AV= 2VN
BTL

=:

'0

z

RL=40
AV=-2VN
BTL

·z1

+

+
c

c
o

0

'E

'E

*'

i

iQ

'c0

14Ji~

]i

0.1

'7

"~

..
r--.

....

z

d

i

~~

Po = 0.25 W

J:

]i 0.1%

PO= 1.0

wf=:
r---

jIP"

j!:

0.01%
20

l""- I'-

~

-

I

Z

+
Q

f=20Hz

10k 20k

m

0.01%
0.01

0.1
Po - Output Power -

f - Frequency - Hz

Figure 9

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

OUTPUT POWER

10%

10%

•r!!

z0

r- i"""

+
c

~

..

f=15kHz

1"'"-1"""

1%

S

-

I

0

I'-r-

J:

iii 0.1%

~I

Z

+
Q

I

1%

RL=40
AV =-6 VN
BTL

J:

0.01%
0.01

f=1kHz
I IIII

'1'""+-1.
~,.....

nt±fjj ~

"'"

I-

I

~ldol~~

0.1%

f=20Hz
II
RL=40
AV=-12VN
BTL

I I 11111
0.1
Po - Output Power -

~
0.01%
0.01

w

Figure 11

11111111
0.1
Po - Output Power - W

Figure 12

~TEXAS

INSTRUMENTS
3-614

I

fJ
f=1kHz

c

i

f= 15 kHz

I'-

,

is
.!:!

10

w

Figure 10

TOTAL HARMONIC DISTORTION PLUS NOISE

..

/

r--J IIIIU

J:

1k

f=1kHz
I IIIII

I

I-

100

1

f=15kHz

..

.~

i

1%

0

PO=1.5W

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

10

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY
10%

10%

Iz

,

f= 15 kHz

....

+

~

;

1%

_

6

~0

Iz

I

-

I

+

i

f=1kHz

~

i

1

--

c

f=20Hz

~

PO=0.25W

u

~

0.1%

RL=4n
Ay =-24 YN
t- BTL
j lB1llll
0.01%
0.01
0.1
Po - Output Power - W

..... I--'

z

PO=0.5W

~

J:

.....

II IIII

0.001%
20

10

100

lk

Figure 14

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

II

10%

PO=lW
RL=8n
BTL

+

lJJllill 1

1%

10k 20k

f - Frequency - Hz

Figure 13

I

~ ~~
PO=1.0W -

!
If 0.01%

I

+
Q

1%

is

r-r-.I-o

0.1%

RL=8n
Ay= 2YN
BTL

~

Ay=-24YN

RL=8n
Ay= 2YN
BTL

2l
"0
z

+

6

1%

f=15kHz

.~ 0.1%

fl= 11 k~ I

'E

I-

.!:!

0.1%

Ay=-12YN

/'

i

.;'

Q

:/1/

l1ll-"'-

u

Ay= 2YN ::

JIf 0.01 .,

~

Ay=~YN

r--..

r--..

f=20Hz

r--..

z

~

~

~
0.001%
20

i

!If 0.01%

I

~
lk

100

10k 201

0.001%
0.01

f - Frequency - Hz

Figure 15

0.1
Po - Output Power - W

10

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

:HI15

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
va
OUTPUT POWER
10%

TOTAL HARMONIC DISTORTION PLUS NOISE
va
OUTPUT POWER
10%

i==

RL=SO
~ Ay=~YN
r-- BTL

-

-

I
f= 15 kHz

1'-:--

II

1%

--

1= 15 kHz

r-I":--

~

1=1 kHz
-.L I III

1=1 kHz

t-r-

-.J.J 1111

If}~

:---1""

0.1%

1= 20 Hz

.:::"'!.

0.01%
0.01

rr-

RL=SO
Ay::i-12YN
BTL

0.01%
0.01

0.1
Po - Output Power - W

Figure 17

Figure 18

TOTAL HARMONIC DISTORTION PLUS NOISE
va
OUTPUT POWER

TOTAL HARMONIC DISTORTION PLUS NOISE
va
FREQUENCY
10%

10%

J

--

i

+

c

~

a
I!

1%

i":--

I

I~

~

+

~

,..

I

0.1%

J

t~JoI~

~

0.1%

1=

If
0.01%
z

~~

III

~

lllUI

l.oiii

PO=JJ~w

§

i!:
III1

0.1
1
Po - Output Power - W

10

100

1k
f - Frequency - Hz

Figure 20

Figure 19

~TEXAS

3-616

PO=75mW

~

RL=SO
t- Ay=-24YN
BTL

0.01%
0.01

PO=25mW 11111 JIIIl

I

r-

i!:

1%

j

I

CI

RL=320
Ay=-1 YN
SE

c

1=1 kHz

.......

c0

{!

J II II

1= 15kHz

CI

.2

10

0.1
Po - output Power - W

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

10k 20k

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETIINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

FREQUENCY

10%

10%
RL=320
Ay=-1 YN
SE

I
+

II
Iz
I

RL=10kO
Ay= 1 YN
SE

::

~

+

t

1%

is

I - - f=15kHz

.2

15

0.1%

!:!i

;-

j!:

0.1%

I
I

I=- f=1 kHz
0.01%

1%

f=20HZ~

YO=1 YRMS

If 0.01%
z
!:!i
l:

~

I-

0.001%
0.01

0.001%
20

0.1
Po - Output Power - W

100

Figure 21

OUTPUT NOISE VOLTAGE

vs

vs

OUTPUT VOLTAGE

BANDWIDTH

10%

100

RL=10kO
Ay=-1 YN
SE

~

LL

>:i

1%

I

~

\

'"

!

~

-r--

-

f = 15 kHz

.~

Ay =-24 YN
D.

60

"0

z

50

:i
a.
:i

40

0

~

j!:

I

Ay~llltj

0.2 OA 0.6 0.8

1

1.2

30 r--

Ay=-6YN ..... 10-

c

>

f=1 kHz

o

J

70

CD

III

f= 20 Hz

If 0.01%

0.001%

80

I
CD

0.1%

Y~DI=5V

90 t-R =40

1

+

II

10k 20k

Figure 22

TOTAL HARMONIC DISTORTION PLUS NOISE

.I

1k
f - Frequency - Hz

20
10

1.4 1.6 1.8

2

o

.....

".

---

10

YO - Output Yoltege - YRMS

Figure 23

...

Ily
.oW

1\
II

K
~

V
V

V

f'1'""'"

tIE
Ay =-2 YN

100
1k
BW • Bandwidth· Hz

10k

Figure 24

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-617

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO

SUPPLY RIPPLE REJECTION RATIO

vs

vs

FREQUENCY

FREQUENCY

o

0

III
'a
I
0

RL=8Q
CB=0.47 I1F
BTL

-20

!g

-20

r--r-.

I

ic

j

-40

.2

J

-60

.!!!

a.

~

-80

-40

..........

c
o

;-..
\~

AV=-24VN

"- ~

~

a.
a.
~
rn -100

IAvl=

V

-

I-

i'-r-.

AV=-1 VN

V

~

-60

'-

{
-80

i

~ -100

Ii ~Til

-120
20

10k 20k

1k

100

I
~

t - ~-

.....

-120
20

RL=32Q
CB=0.47 I1F
SE

1k

100

10k 20k

f - Frequency - Hz

f - Frequency - Hz

Figure 25

Figure 26

CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

0
PO=1W
RL=8Q
Av =-2 VN
BTL

-20

-20

-40

III
'a
I

...

..e

li

III
'a
I

-60

-100

-120

-40~~+#~-r~~~--~~+#~~

1

-60

/'"

0

-80

PO=1W
RL=8Q
Ay=-24VN
BTL

-

-

20

LEFT TO RIGHT
..........

RI~H~ ~6ll~~

~IJ...~

100

.,,'"
~

1k

o

./

LEFT TO RIGHT

-100

r---

....

T~ ~EW+H+-/--+-+-++++H+---l

~~

-120L-J...."L",LI...I..I.I..I.l-IIIII-'--!-IIIL...L..U.J..I,LII
20
100
1k

f - Frequency - Hz

f - Frequency - Hz

Figure 28

Figure 27

-!!1TEXAS

INSTRUMENTS

3-618

RIGHT

...
10k 20k

..... ~

-80~~~~~~~~~-r
.J,.. JJi
.,,~

POST OFFICE BOX 655303 • DAlLAS. TEXAS 75265

10k 20k

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
CROSSTALK

SHUTDOWN ATTENUAllON

vs

va

FREQUENCY

FREQUENCY

0

-20

0
VO=1 VRMS
RL=10kn
Ay=-1 VN
SE

VI=1 VRMS
RL= 10 kO, SE

-40

III

III

"I

"I

1

111111

-20

-40

c
0

-60

i

~

-60
RL=32o,SE

c

Ii!
(J

LEFT TO RIGHT

-80

~

-

-80

i"'-o.~
-100

-100
RIGHT TO LEFT

-120
20

I I I 111111
100

1k
f - Frequency - Hz

-120
20

10k 20k

100

Figure 29

tj8~i(

1k
f - Frequency - Hz

10k 20k

Figure 30
SIGNAL-TQ-NOISE RAllO

vs
120
115
III

"0
I

Ij
~

..
~
c

110
105

r
a:

90

z

,

"AV=11

~~

t--.....

....

t:--..

'

PO=1W
RL=8n
BTL

UIII

...

1111
Av =-12 VN

r-

~

i'-..~ ...

~

t-- r...

100 --" AV =-2 VN

95

I

BANDWIDTH

--~

~ r...

AV =-6 VN

II)

85
80
20

100

1k
BW - Bendwldth - Hz

...........

:---....

"
10k 20k

Figure 31

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, 1EXAS 75265

3-619

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE
1800

10

Illtil

7.5
5

1/

900

c

2.5
ID

"c
I

'iii
Cl

..

.~

~

::E

Phase

0

.8:

00

~

I'.

-2.5

D.
I

....E

RL=8Q
AV=-2VN
BTL

-5

111111111

-7.5
-10
10

-900

1111,

1111111111
100

1k

10k

100k

1M

_180 0
2M

f - Frequency - Hz

Figure 32

CLOSED LOOP RESPONSE
30

1800

II~UI

25

r-...

20
15

~

c

'e»

ID

"c
I

'iii
Cl

900

10

~

Phase

i'

~

5
0
-5

-10
10

RL=8Q
AV=-6VN
BTL
111111111

~

1k

10k

100k

1M

f - Frequency - Hz

Figure 33

~TEXAS

INSTRUMENTS
&-620

....E
-900

11I11111111
100

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

~

D.
I

1\

1111

II

00

2M

-180

0

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE
30

11~1

11111

RL=80
AV =-12 VN
BTL

25

90°

20

Gain
ID

15

'D

~

~

10

c

.~

11~~a~

V

01

:::E
0°

3l01

s::.

11-

"-

5

I

E
-e-

o

-900

-10
10

~
~\
100

1k

10k

100k

1M

-180°
2M

f - Frequency - Hz

Figure 34

CLOSED LOOP RESPONSE
30

8'0

RL =
AV =-24 VN

111111111

G~'~""-

25
20

ID

15

90°

II

'D

~

~

~TL

10

c

'E'
01
i'o

Phase

0°

t"-

01
s::.

11I

5

E

i\

o

-10
10

..

==GI

-e-SOO

l\,

-180°
100

1k

10k

100k

1M

2M

f - Frequency - Hz

Figure 35

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

3-621

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SEmNGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

va

,

3.5
3
~

.

II..

::I

0

2.5

1500

Ay=-2VN

AV=-1 VN

BTL

SE

~I

1\

,

2

.
I

10%THD+N

.\

1.5

1

~

I

rP

8

R

1%THD~~ r--..

0.5

o

LOAD RESISTANCE

1250

I

I
I

va

LOAD RESISTANCE

o

8

~

N

1000

750

~ 10%THD+N

500

\~

250

U

~

M

~

o

M

1%TH~~ r....
o

I

I

8

~

RL - Load Resistance - 0

N

Figure 36

OUTPUT POWER

1.8

I

i

1.2

I

0.8

~

II..
I

0.6

Q

II..

0.4

-

~

/

//
II
V
r/

0.35
~

---

.....I'"

I

0

I

0.2

-'-

0.15

I--

r--....

I

8~

~

1.5

2

Po - Output Power -

w

0.1

2.5

o

o

........

~

~

1=1 kHz

1

~

SE
Each Channel
~

U

~

Po - Output Power -

Figure 38

Figure 39

:'IlExAs
INSTRUMENTS
POST OFFICE BOX 855303 • DAllAS. TEXAS 75265

40-

K

1"",,80
to....

0.05 ~ UO

BTL
Each Channel
1

V """"

1

J V'L

1=1 kHz

0.5

0.25

r--.

;/

0.3

c:

40

0.2

o
o

0.4

30 -

/'

1A

M

va

OUTPUT POWER

c:

M

~

POWER DISSIPAnON

va

~

~

Figure 37

POWER DlSSIPAnON

1.6

U

RL - Load Resistance - 0

U

w

V

U

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S285 - NOVEMBER 1999

APPLICATION INFORMATION
POWER DISSIPATION
vs
AMBIENT TEMPERATURE
7

\

ElJA4

6
~
I

c

5

i

4

0

0;

~

i

0
D.
I
Q

D.

3

I I
ElJA1,2

2

1\

"- ~

ElJA3

ElJA1 =45.9°CIW
ElJA2 = 4S.2°CIW
ElJA3 =31.2°CIW
ElJA4 =18.6°CIW

'""" ~

\

""',
\

1\
\
~
.......

o

~~O

~~

0
~
~
~
00 1001~1~1~
TA - Ambient Temperature - °C

Figure 40

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3--623

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 41)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine~pitch, surface-mount package can
be reliably achieved.

Side View (a)

Thermal

Pad

End View (b)

Bottom View (c)

Figure 41. Views of Thermally Enhanced PWP Package

Figure 42 and Figure 43 are schematic diagrams of typical notebook computer application circuits.

~TEXAS

INSTRUMENTS
3-624

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285-NOVEMBER 1999

Right CIRHP
Head- OA71lF
phone
Input
Signal
20

-1

CIRLINE
Right 0.471lf
Line
Input
Signal

23

RHPIN
RLiNEIN

-1

a
CRIN
0.471lF

ROUT+

21

ROUT-

16

RIN

T

-=PCBEE~ 14

PC-BEEP
Input ~I----'.:!..f--"-""::!!!'!::!'""--I
Signal CPeB
0.471lF

100kn

PVDD

1a SaaNotaA

I-.!...!.=f-'~-'--

Dapop
Circuitry

Left CILHP
Haad- 0.47 IlF..,..,Y-=-:.==--_--'
phone
Input
Signal

Po_r
Management

---1

VDD

19

BYPASS
SHUT-

11

-:J'
-

VDD

'I'

I-D=.;O::.;W:..:.;N:.:..t--,,2=:.2....,

I~t----"M.-.......~w;::::::::;-Tr--'G=N,:,Dll

VDD

CSR
0.1 !If

-=-C

CSR
0.11lF

BVP

-:J'

LOUT+

To
0.47 11F
SystamControl
1,12,
4
13,24

LOUT-

9

1 kG

LIN
CLiN
0.471lF

100kn
NOTE A.

A 0.1 IlF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency nOise signals, a larger
electrolytic capacitor of 10 IlF or greater should be placed near the audio power amplifier.

Figure 42. Typical TPA0222 Application Circuit USing Single-Ended Inputs and Input MUX

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-625

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

APPLICATION INFORMATION
N/C
20
CIRINRight 0.47 IlF
23
Negative ~
Differential
Input
Signal

CIRIN+
Right 0.47 IlF
8
Positive
Differential
Input
Signal

-1

PC BEEP
14
Input
Signal CpCB
0.47 1lF

---1

2
3

RHPIN
RLINEIN

R
MUX
ROUT+

21

ROUT-

16

PVDD

18

VDD

19

BYPASS

11

RIN

P~EEP§

Beep·

GAINO
GAIN1
SElBTL

See Note A

VDD
CSR
-:J:'0.1IlF
VDD

I-~==+--=----e--

Galnl
MUX
Control

Depop
Circuitry
Power
Management

HP/LINE
N/C

'I'

CSR
0.11lF

22
LHPIN

GND
L
MUX

LLiNEIN
Left
CILlNNegative 0.47 IlF
Differential ~
Input
Signal
CILlN+
Left 0.471lF
Positive -.'l
10
LIN
Differential 7r--+=-t-='-'-----~..
Input
Signal

CBYP
To -:J:' 0.47 IlF
SystemControl

LOUT+

LOUT-

1 kO

COUTL
330IlF

9

100kO

NOTE A.

A 0.1 IlF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals. a larger
electrolytic capacitor of 10 IlF or greater should be placed near the audio power amplifier.

Figure 43. Typical TPA0222 Application Circuit Using Differential Inputs

~TEXAS

INSTRUMENTS
3--626

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

APPLICATION INFORMATION
gain setting via GAINO and GAIN1 inputs
The gain of the TPA0222 is set by two input terminals, GAINO and GAIN1.

Table 1. Gain Settings
GAINO

GAIN1

SE/BTL

Av

0

0

0

0

1

0

1

0

0

1

1

0

X

X

1

-2VN
-6VN
-12VN
-24VN
-1 VN

The gains listed in Table 1 are realized by changing the taps on the input resistors inside the amplifier. This
causes the input impedance, ZI, to be dependant on the gain setting. The actual gain settings are controlled
by ratios of resistors, so the actual gain distribution from part-to-part is quite good. However, the input
impedance will shift by 30% due to shifts in the actual resistance of the input impedance.
For design purposes, the input network (discussed in the next section) should be designed assuming an input
impedance of 10 kil, which is the absolute minimum input impedance of the TPA0222. At the higher gain
settings, the input impedance could increase as high as 115 kil.

input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r--------------------'IICf-~~.!:.N~III-~1
.
Slgnal~ 1
I

ZF

Input

"1 I
I

-=-

The typical input resistance at each gain setting is given in the table below:

Av

ZI

-24VN
-12VN
-6VN
-2VN

14 kil

26 kil
45.5 kil
91 kn

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-627

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL

SLOS285-NOVEMBER 1999

APPLICATION INFORMATION
The -3 dB frequency can be calculated using equation 1:

f

-3

-

1

dB. - 23t C(R II R,)

(1 )

If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addition, the order of the filter could be increased.
Input capacitor, C,
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input Signal to the
proper dc level for optimum operation. In this case, C, and the input impedance of the amplifier, Z" form a
high-pass filter with the comer frequency determined in equation 2.

tC(highpass) =

(2)

2n~,c,

The value of C, is important to consider as it directly affects the bass (lOW frequency) performance of the circuit.
Consider the example where Z, is 710 kn and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
C 1
, - 2nZ, tc

(3)

In this example, C, is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 I1F. A further
consideration for this capacitor is the leakage path from the input source through the input network (C,) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capaCitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.

3-628

:illExAs
INSTRUMENTS
POST OFFICE BOX 656303 • DALLAS, TEXAS 75285

TPA0222
STEREO 2-W AUDIO POWER AMP'LIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285-NOVEMBER 1999

APPLICATION INFORMATION
power supply decoupling, Cs

The TPA0222 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 I1F placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capaCitor of 10 I1F or greater placed near
the audio power amplifier is recommended.
midrall bypass capacitor, CBYP

The midrail bypass capacitor, CSyp. is the most critical capaCitor and serves several important functions. During
start-up or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the midrail generation circuit internal to the amplifier, which appears as degraded PSRR and
THD+N.
Bypass capacitor, CSyp. values of 0.4711F to 111F ceramic or tantalum low-ESR capacitors are recommended
for the best THO and noise performance.

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cd is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capaCitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

fC(hlgh)

=

23t~L Cc

(4)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 I1F is chosen and loads vary from 3 n,
4 n. 8 n, 32 n. 10 kil, to 47 kil. Table 2 summarizes the frequency response characteristics of each
configuration.

-!II

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-629

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL

SLOS285 - NOVEMBER 1999

APPLICATION INFORMATION
Table 2. Common Load Impedances Vs Low Frequency Output Characteristics In SE Mode
RL

Cc

Lowest Frequency

311

330llF

161 Hz

411

3301lF

120 Hz
60 Hz

811

330llF

3211

330llF

15 Hz

10,00011

330llF

0.05 Hz

47,00011

330llF

0.01 Hz

As Table 2 indicates, most of the bass response is attenuated into a 4-n load, an 8-n load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

bridged-tied load versus single-ended mode
Figure 44 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0222 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but initially consider power to the load. The differential drive to the speaker
means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the
voltage swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power
equation, where voltage is squared, yields 4x the output power from the same supply rail and load impedance
(see equation 5).

v

_ VO(PP)

(nns) -

(5)

2/2

2
V(nns)

Power = - RL

~TEXAS

INSTRUMENTS
3-630

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285-NOVEMBER 1999

APPLICATION INFORMATION
VDD

*
J'!

J'
RL

VO(PP)

2x vO(PP)

'V

*

-vO(PP)

Figure 44. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an B-n speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvementwhich is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 45. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 331lF to 1000 IlF)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
fc =

(6)

1

21tRL C c

For example, a 68-IlF capacitor with an B-n speaker would attenuate low frequencies belOW 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD

~dB~-----J~=====

Figure 45. Single-Ended Configuration and Frequency Response

~TEXAS

INSTRUMENTS
POST OFFICE BOX 65S303 • DAUAS. TEXAS 75285

3-631

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH ·FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285- NOVEMBER 1999

APPLICATION INFORMATION
Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

single-ended operation
In SE mode (see Figure 44 and Figure 45), the load is driven from the primary amplifier output for each channel
(OUT+, terminalsi 21 and 4).
The amplifier switches single-ended operation when the SE/BTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain to 1 VN.

input MUX operation
The input MUX allows two separate inputs to be applied to the amplifier. This allows the designer to choose
which input is active independent of the state of the SE/BTL terminal. When the HP/LINE terminal is held high,
the headphone inputs are active. When the HP/LINE terminal is held low, the line BTL inputs are active.

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
VDO. The internal voltage drop multiplied by the RMS value of the supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 46).
100

/

--rvvvvffll.-

V(LRMS)

IOO(avg)

Figure 46. Voltage and Current Waveforms for BTL Amplifiers
Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it isa full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.

~TEXAS

INSTRUMENTS
3--632

POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETIINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

APPLICATION INFORMATION
Efficiency of a BTL amplifier =

p--'P=--

(7)

SUP

Where:
VLrms2
Vp
V 2
PL = - R - ' andV LRMS - therefore, PL = ~
L
- 12'
2RL

" VRP sin(t) dt = ~

= ~J
o

looavg

and

L

V
1t
x RP [cos(t)] 0
L

=

2V
1t :

L

Therefore,

_ 2 Voo Vp
Psup -

11:

RL

substituting PL and PSUP into equation 7,
Vp2

Efficiency of a BTL amplifier
Where:

21\

PL =Power devilered to load
PSUP = Power drawn from power supply
VLRMS =RMS voltage on BTL load
RL = Load resistance
V P = Peak voltage on BTL load
looavg =Average current drawn from
the power supply
Voo =Power supply voltage
llBTL = Efficiency of a BTL amplifier

Vp
2 V DO V P = 4 V DO
11: RL
1t

Therefore,

_1t~
IlBTL -

(8)

4 Voo

Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.

Table 3. Efficiency Vs Output Power in 5-V 8-0 BTL Systems
Output Power

Efficiency

Peak Voltage

(W)

(%)

(V)

Internal Dissipation

0.25
31.4
2.00
0.50
44.4
2.83
1.00
62.8
4.00
4.47t
1.25
70.2
t High peak voltages cause the THO to increase.

(W)

0.55
0.62
0.59
0.53

A final pOint to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, VDD is in the denominator. This
indicates that as VDD goes down, efficiency goes up.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-633

TPA0222
STEREO 2-WAUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285-NOVEMBER 1999

APPLICATION INFORMATION
crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the TPA0222 data sheet, one can see
that when the TPA0222 is operating from a 5-V supply into a 3-n speaker 4-W peaks are available. Converting
watts to dB:
PdB

P

= 10Log~ =
P ref

10Log 4
1 Ww

=

6 dB

(9)

Subtracting the headroom restriction to obtain the average listening level without distortion yields:

6 dB -15 dB = -9 dB (15 dB crest factor)
6 dB -12 dB = ~ dB (12 dB crest factor)
6 dB - 9 dB = -3 dB (9 dB crest factor)
6 dB - 6 dB
6 dB - 3 dB

=0 dB (6 dB crest factor)
=3 dB (3 dB crest factor)

Converting dB back into watts:
Pw = 10PdBj10 x Pref

(10)

= 63 mW (18 dB crest factor)
= 125 mW (15 dB crest factor)
= 250 mW (9 dB crest factor)

= 500 mW (6 dB crest factor)
= 1000 mW (3 dB crest factor)

= 2000 mW (15 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for as-V, 3-n system, the internal dissipation in the TPA0222 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0222 Power Rating, 5-V, 3-0., Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/ehennel)

MAXIMUM AMBIENT
TEMPERATURE
':'3°C

4

2W(3dB)

1.7

4

1000 mW (6 dB)

1.6

6°C

4

500 mW (9 dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.6

78°C

4

63 mW (18 dB)

0.6

96°C

:lllExAsINSTRUMENTS
POST OFFICE BOX 855303 • DAllAS. TEXAS 75285

TPA0222
STEREO 2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

APPLICATION INFORMATION

crest factor and thermal considerations (continued)
Table 5. TPA0222 Power Rating, 5-V,

a-a, Stereo

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

2.5W

1250 mW (3 dB crest factor)

0.55

100°C

2.5W

1000 mW (4 dB crest factor)

0.62

94°C

2.5W

500 mW (7 dB crest factor)

0.59

97°C

2.5W

250 mW (10 dB crest factor)

0.53

102°C

PEAK OUTPUT POWER

The maximum dissipated power, POmax , is reached at a much lower output power level for an 8-0 load than for
a 3-0 load. As a result, this simple formula for calculating POmax may be used for an 8-0 application:

POmax

2Vfm
= n;2R

(11)

L

However, in the case of a 3-0 load, the POmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 0 load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to 8JA:

e

JA

=

1

Derating Factor

= _1_
0.022

= 450C/W

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given 8JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0222 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
TA Max = T J Max - e JA Po

=

150 - 45(0.6 x 2)

(13)

=

96°C (15 dB crest factor)
NOTE:

Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.
TableS 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0222 is deSigned with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-0 speakers dramatically increases the thermal performance by increasing amplifier
efficiency.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

~35

TPA0222
STEREO'2-W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SLOS285 - NOVEMBER 1999

APPLICATION INFORMATION
SE/BTL operation
The ability of the TPA0222 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0222, two separate amplifiers drive OUT+ and OUT-. The SE/BTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When
SE/BTL is held low, the amplifier is on and the TPA0222 is in the BTL mode. When SElBTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0222 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SElBTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 47.

20

RHPIN

23

RLINEIN

R

MUX
ROUT+

8

21

RIN

VDD
ROUT-

16

l00kQ
SEieTL

15 100 kG

~

n

.-----~
Figure 47. TPA0222 Resistor Divider Network Circuit

Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-k.Q/1-kQ divider pulls the SE/BTL input low. When a plug is inserted, the 1-kQ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT-amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.

~TEXAS

INSTRUMENTS
3-636

POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

TPA0222
STEREO 2·W AUDIO POWER AMPLIFIER
WITH FOUR SELECTABLE GAIN SETTINGS AND MUX CONTROL
SL0S285 - NOVEMBER 1999

APPLICATION INFORMATION

PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is activated automatically. When the PC BEEP input is active,
both of the LlNEIN and HPIN inputs are deselected and both the left and right channels are driven in BTL mode
with the signal from PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is
independent of the volume setting. When the PC BEEP input is deselected, the amplifier will return to the
previous operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC
BEEP will take the device out of shutdown and output the PC BEEP Signal, then return the amplifier to shutdown
mode.
The preferred input signal is a square wave or pulse train with an amplitude of 1 Vpp or greater. To be accurately
detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall times of less than 0.1 ~ and a
minimum of 8 riSing edges. When the Signal is no longer detected, the amplifier will return to its previous
operating mode and volume setting.
If it is desired to ac-couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
equation 14:
C

>

PCB - 211:

f pCB1(100 kQ)

(14)

The PC BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.

shutdown modes
The TPA0222 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, Ipp = 150 !lA. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.
Table 6. HP/LlNE, SE/BTL, and Shutdown Functions
AMPLIFIER STATE

INPUTst
HPILINE

SElBTL

SHUTDOWN

INPUT

OUTPUT

X

X

Low

X

Mute

Low

Low

High

Line

BTL

Low

High

High

Line

SE

High

Low

High

HP

BTL

High

High

High

HP

SE

t Inputs should never be left unconnected.
X do not care

=

~TEXAS

INSTRUMENTS
POST OFF1CE BOX 655303 • DALLAS, TEXAS 75265

3-637

TPA0223
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
OOQPACKAGE
(TOP VIEW)

• Ideal for Notebook Computers, PDAs, and
Other Small Portable Audio Devices
• 2 W Into 4-0 From S-Y Supply
• 0.6 W Into 4-0 From 3-Y Supply

MONO-IN
SHUTDOWN

• Stereo Head Phone Drive
• Separate Inputs for the Mono (BTL) Signal
and Stereo (SE) Left/Right Signals

Vee
BYPASS
RIN

LOIMO
LIN
GND

SRIMN
ROIMO

• Wide Power Supply Compatibility
3YtoSY
• Meets PC99 Desktop Specs (Target)
• Low Supply Current
- 11 mA Typical at S Y
- 10 mA Typical at 3 Y
• Shutdown Control ••• 1 IlA Typical
• Shutdown Pin Is TTL Compatible
• -40°C to 8SoC Operating Temperature
Range
• Space-Saving, Thermally-Enhanced MSOP
Packaging

description
The TPA0223 is a 2-W mono bridge-tied-Ioad (BTL) amplifier designed to drive speakers with as low as 4-0
impedance. The amplifier can be reconfigured on-the-fly to drive two stereo single-ended (SE) signals into head
phones. This makes the device ideal for use in small notebook computers, PDAs, Digital Personal Audio
players, anyplace a mono speaker and stereo head phones are required. From a 5-V supply, the TPA0223 can
delivery 2-W of power into a 4-0 speaker.
The gain of the input stage is set by the user-selected input resistor and a 50-kn internal feedback resistor
(Av =- RF/ RI)' The power stage is internally configured with a gain of -1.25 VN in SE mode, and -2.5 VN in
BTL mode. Thus, the overall gain of the amplifier is 62.5 kn/ RI in SE mode and 125 knt RI in BTL mode. The
input terminals are high-impedance CMOS inputs, and can be used as summing nodes.
The TPA0223 is available in the 10-pin thermally-enhanced MSOP package (DGQ) and operates over an
ambient temperature range of -40°C to 85°C .

.A.

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments InCOrporated.

~TEXAS

Copyright © 2000, Texas Instruments Incorporated

INSTRUMENTS
POST OFFICE BOX 665303 • DALLAS, TEXAS 75265

3-639

TPA0223
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS277A - JANUARY 2000 - REVISED MARCH 2000

VDD

VDD

BYPASS
Mono
Audio

CI

Input

!I--

Right
Audio

CI

Input

!I----'\N\r---'

50110
J

V\lv---,

,
,,
,,

BYPASS

,

so 110
Stereo/Mono
Control

so 110
Left
Audio
Input

,
,
,

1.2S*R

,I
,
,

CI

9 LIN
!~~R~I~--~----1

From
System Control

Cc

LOlMo- '10

,
,
,
,
,

BYPASS

2' SHUTDOWN

-=-

STIMN ' 7

Shutdown
and Depop
Circuitry

L _________________________

,
,
,
,
,
,
,
,
,

~

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

MSOpt
(DGO)

-40°C to 85°C

TPA0223DGQ

MSOP
SYMBOLIZATION
AEI

t The DGQ package are available taped and reeled. To order a taped and reeled part. add the
suffix R to the part number (e.g., TPA0223DGQR).

~TEXAS

3-640

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

1110

TPA0223
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS277A - JANUARY 2000 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME

110

NO.

DESCRIPTION

MONO·IN

1

I

SHUTDOWN

2

I

SHUTDOWN places the entire device in shutdown mode when held low. TTL compatible input.

VDD

3

I

VDD is the supply voltage terminal.

BYPASS

4

I

BYPASS is the tap to the voltage divider for internal mid·supply bias. This terminal should be connected
to a O.l·!1F to l-!1F capacitor.
Right·channel input terminal

Mono input terminal

RIN

5

I

ROIMO

6

0

Right-output in SE mode and mono positive output in BTL mode

SRIMN

7

I

Selects between stereo and mono mode. When held high, the amplifier is in SE stereo mode, while held
low, the amplifier is in BTL mono mode.

GND

8

LIN

9

I

Ground terminal

LOIMO

10

0

Left-channel input terminal
Left-output in SE mode and mono negative output in BTL mode.

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/t6 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
DGQ

DERATING FACTOR
2.14 w1I

17.1 mWrC

TA=85°C
1.37W

1.11 W

'11 Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002), for more information on the PowerPAD package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VDD
High-level input voltage, V,H

STIMN

IVDO =3V
IVDD=5V

SHUTDOWN
Low-level input voltage, V,L

STIMN

MIN

MAX

2.5

5.5

UNIT
V

2.7
4.5

V

2

IVOD=3V

1.65

IVDO=5V

2.75

SHUTDOWN
-40

Operating free-air temperature, TA

V

0.8
85

°c

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-641

TPA0223
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS277A - JANUARY 2000 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, Voo = 3 V, TA ;: 25°C (unless otherwise
noted)
PARAMETER

TEST CONomONS

IVOOI

Output offset voltage (measured differentially)

100

Supply current

IDD(SD)

Supply current, shutdown mode

operating characteristics,

MIN

TYP

MAX

UNIT

30

mV

10

13

mA

1

10

IiA

TYP

MAX

Voo =3 V, TA =25°C, RL =4 n

PARAMETER

TEST CONomoNS
THO = 1%,

BTL mode

THD=0.1%,

SEmode,

Po

Output power, see Note 1

THO+N

Totel harmonic distortion plus noise

Po =500 mW,

1=20 Hz to 20 kHz

BaM

Maximum output power bandwidth

Gain=2,

THO=2%

MIN

660

mW

33

RL=320

UNIT

0.3%
20

kHz

NOTE 1: Output power is measured at the output terminals 01 the device at 1 = 1 kHz.

electrical characteristics at specified free-air temperature, Voo
noted)
PARAMETER

TEST CONOmONS

IVOOI

Output offset voltage (measured differentially)

100

Supply current

IDO(SD)

Supply current, shutdown mode

operating characteristics, Voo

=5 V, TA =25°C (unless otherwise
MIN

TYP

MAX

UNIT

30

mV

11

15

mA

1

10

IiA

=5 V, TA:: 25°C, RL =4 n

PARAMETER

TEST CONomONS
THO = 1%,

BTL mode

THO = 0.1%,

SEmode,

Po

Output power, see Note 1

THD+N

Total harmonic distortion plus
nOise

PO=1 W,

1=20 Hz to 20 kHz

BOM

Maximum output power bandwidth

Gain =2.5,

THO=2%

MIN

RL=320

NOTE 1: Output power is measured at the output terminals 01 the device at 1 = 1 kHz.

~TEXAS

INSTRUMENTS

POST OFFICE aox 655303 • DALLAS. TEXAS 75265

TYP

MAX

UNIT

2

W

95

mW

0.2"10
20

kHz

TPA0232
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
• Compatible With PC 99 Desktop Line-Out
Into 10-ka Load
• Compatible With PC 99 Portable Into 8-n
Load
• Internal Gain Control, Which Eliminates
External Gain~Setting Resistors
• DC Volume Control From +20 dB to -40 dB
• 2-W/Ch Output Power Into 3-n Load
• Input MUX Select Terminal
• PC-Beep Input
• Depop Circuitry
• Stereo Input MUX
• Fully Differential Input
• Low Supply Current and Shutdown Current
• Surface-Mount Power Packaging
24-Pin TSSOP PowerPADTM

PWPPACKAGE
(TOP VIEW)

GND
HPILINE
VOLUME
lOUT+
lLiNEIN
lHPIN
PVoo
RIN
lOUTLIN
BYPASS
GND

10
2
3
4
5

6
7
8
9
10
11
12

24
23
22

21
20
19
18
17
16
15
14
13

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
Voo
PVoo
ClK
ROUTSE/BTl
PC-BEEP
GND

description
The TPA0232 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-n loads. This device minimizes the number of
external components needed, which simplifies the design and frees up board space for other features. When
driving 1 W into 8-n speakers, the TPA0232 has less than 0.4% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
Amplifier gain is controlled by means of a dc voltage input on the VOLUME terminal. There are 31 discrete steps
covering the range of +20 dB (maximum volume setting) to -40 dB (minimum volume setting) in 2 dB steps.
When the VOLUME terminal exceeds 3.54 V, the device is muted. An internal input MUX allows two sets of
stereo inputs to the amplifier. The HP/LINE terminal allows the user to select which MUX input is active
regardless of whether the amplifier is in SE or BTL mode. In notebook applications, where internal speakers
are driven as BTL and the line outputs (often headphone drive) are required to be SE, the TPA0232
automatically switches into SE mode when the SElBTL input is activated, and this effectively reduces the gain
by6dB.
The TPA0232 consumes only 10 mA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to less than 150 J.LA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously aChievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are readily realized in multilayer PCB
applications. This allows the TPA0232 to operate at full power into 8-n loads at ambient temperatures of 85°C.

•.

~

Please be aware that an important notice conceming availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

Copyright © 1999, Texas Instruments Incorporated

3-643

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS288-NOVEMBER 1999

functional block diagram

RH~N

~

RUNEIN _ _....,

M~X

""--...,---'

>-......- - - - - -

ROUT+

VOLUME - - - - - - -..

RIN

--------t---+--e

ROUT-

PCOBEEP--1

PC

Beep
Power

Management
SeJBTL==1
MUX
Control
HPILINE

LHPIN
LLiNEIN

PVDD
VDD
BYPASS
SHUTDOWN
GND

(;gLOUT+

LlN---------~

___

>-_____- - - - - -

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75285

LOUT-

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286 - NOVEMBER 1999

AVAILABLE OPTIONS
PACKAGED DEVICE
TA

TSSOpt
(PWP)

-40°C to B5°C

TPA0232PWP

t The PWP package IS available taped and reeled. To order a taped and reeled part,
add the suffix R to the part number (e.g., TPA0232PWPR).

Terminal Functions
TERMINAL
NO_
NAME
BYPASS

11

ClK

17

GNO

1,12
13.24

DESCRIPTION

110

Tap to voltage divider for intemal mid-supply bias generator
I

If a 47-nF cepacitor is attached, the TPA0232 generates an intemal clock. An extemal clock cen override
the intemal clock input to this terminal.
Ground connection for circuitry. Connected to thermal pad.

lHPIN

6

I

Left channel headphone input, selected when SElBTl is held high

LIN

10

I

Common left input for fully differential input. AC ground for single-ended inputs.

lllNEIN

5

I

Left channel line negative input, selected when SElBTL is held low

LOUT+

4

Left channel positive output in BTL mode and positive output in SE mode

LOUT-

9

0
0

Left channel negative output in BTL mode and high-impedance in SE mode

HP/LINE

2

I

HPILINE is the input MUX control input. When the HPILINE terminal is held high, the headphone inputs
(LHPIN or RHPIN [6,20]) are active. When the HPILINE terminal is held low, the line BTL inputs (LLINEIN
or RLINEIN [5, 23)) are active.

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > I-V (peak-to-peak) square wave is input
to PC-BEEP.

7,18

I

Power supply for output stage

20

I

Right channel headphone input, selected when SElBTL is held high

PVOO
RHPIN
RIN

8

I

Common right input for fully differential input. AC ground for single-ended inputs.

RLiNEIN

23

I

Right channel line input, selected when SElBTL is held low

ROUT+

21

Right channel positive output in BTL mode and positive output in SE mode

ROUT-

16

0
0

SElBTL

15

I

Hold SElBTL low for BTL mode and hold high for SE mode.
When held low, this terminal places the entire device, except PC-BEEP detect circuitry, in shiJtdown
mode.

Right channel negative output in BTL mode and high-impedance in SE mode

SHUTDOWN

22

I

VOO

19

I

Analog VOO input supply. This terminal needs to be isolated from PVOO to achieve highest performance.

I

VOLUME detects the dc level at the terminal and sets the gain for 31 discrete steps covering a range of
20 dB to -40 dB for dc levels of 0.15 V to 3.54. When the de level is over 3.54 V, the device is muted.

VOLUME

3

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-645

TPA0232
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286-NOVEMBER 1999

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)*
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-airtemperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
=1=

Stresses beyond those listed under "absolute maximum ratings· may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
2.7W§

PWP

DERATING FACTOR

TA=70°C

21.8 mW!OC

1.7W

1.4W

§ Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMAOO2), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
High-level input voltage, VIH

MIN

MAX

4.5

5.5

SElBTl, HP!LlNE

4

SHUTDOWN

2
3

SHUTOOWN

0.8

-40

Operating free-air temperature, TA

V
V

SElBTl, HP!LlNE

lOW-level input voltage, Vil

UNIT

85

V
°C

electrical characteristics at specified free-air temperature, Voo = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

MIN

TYP

MAX

UNIT

IVOOI

Output offset voltage (measured differentially)

VI=O, Ay=2VN

PSRR

Power supply rejection ratio

VOO=4 Vto 5 V

High-level input current

VOO=5.5V,
VI=VOO

.900

nA

IIlll

low-level input current

VOO=5.5V,
VI=OV

900

nA

ZI

Input impedance

IIIHI

100

Supply current

IOO(SO)

Supply current, shutdown mode

67

mV
dB

See Figure 28
BTL mode

10

15

SEmode

5

7.5

150

300

~TEXAS

INSTRUMENTS
3-646

25

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

mA

ItA

TPA0232
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER 1999

operating characteristics, Voo
noted)

=5 V, TA =25°C, RL =4 Q, Gain =2 VN, BTL mode (unless otherwise

PARAMETER

TEST CONDITIONS

Po

Output power

THO=1%,

f= 1 kHz

THO+N

Total harmonic distortion plus noise

PO=1W,

f=20Hzto15kHz

BOM

Maximum output power bandwidth

THO =5%

Supply ripple rejection ratio

f= 1 kHz,
CB=0.47 !LF

Noise output voltage

CB=0.47 !LF,
f = 20 Hz to 20 kHz

Vn

MIN

TYP

MAX

UNIT

2

W

0.4%
kHz

>15
BTL mode

65

SEmode

60

BTL mode

34

SEmode

44

dB

!LVRMS

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power
vsGain

1,4,6,8,10
2

THO+N

Total harmonic distortion plus noise

Vn

Output noise voltage

vs Frequency

13

Supply ripple rejection ratio

vs Frequency

14,15

Crosstalk

vs Frequency

16,17,18

Shutdown attenuation

vs Frequency

19

SNR

Signal-ta-noise ratio

vs Frequency

20

Po

Output power

vs Frequency
vs Output voltage

Power dissipation

ZI

Input impedance

12

21,22

Closed loop response

Po

3,5,7,9,11

vs Load resistance

23,24

vs Output power

25,26

vs Ambient temperature

27

vs Gain

28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-647

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
va
OUTPUT POWER

TOTAL HARMONIC DISTORTION PLUS NOISE
va
VOLTAGE GAIN

10%

1%

Iz

J I

1

+
c

0

'f

1%

~Q

J!
c

RL=4U

=
-

r-

RL=3U

0

=

J

E
!

0.1%

j

/V

1

I ,
+

I /

RL=8U

I- Po = 1 W for Ay~B
~ YO = 1 YRMS for AyS4 dB
I- RL=8U
BTL

J

I

~

J!

~

0.1%

!

j

I

--

I

z

=

0

Ay +2OtoO dB
f= 1 kHz
BTL

j:

0.01%
0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 2.75

~

-

-

j:
0.01%

3

-40

-30

Po - OUtput Power - W

Figure 1

va

FREQUENCY

OUTPUT POWER

10%

10%

RL=3U
Ay = +20toOdB
BTL

·1z

+

+

&

j

Ii

%
PO=1W

J!

J

III
PO=0.5W

~,

0.11'1<

j

1'""

1%

r--

E
!

~

0.1%

~

? I J111W-

j:
0.01 %
20

100

1k
,- Frequency - Hz

11 illt

,=

20 kHz

b

z

~~~

./

'=2OHz

0

RL=3U
Ay=+2OtoOdB
BTL

j:

10k 20k

J
..,

~

I

P"=1.75W'-

0.01%
0.01

Figure 3

0.1
1
Po - Output Power - W

Figure 4

~1ExAs

3-648

20

TOTAL HARMONIC DISTORTION PLUS NOISE

va

I

10

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

j

-20
-10
o
Ay • Yoltage Gain· dB

INSTRUMENTS
POST OFFICE BOX S55303 • DALLAS. TEXAS 75286

10

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10%

10%

+

c

f

1=

RL=4Q
Ay=+20toOdB

r- BTL

+

1%

I

1=

I
g
i

RL=4Q
Ay = +20 to 0 dB
BTL

z=

'0

1%

i"- .... ~

'f=
- 20 kHz

Q

.2
c

C)

0

i
~

PO=O.25W

:c
0.1%

J
I

. / Viii'
PO=1.5W

I

z

~
0.1%

" r--

f=1 kHz

-

If

~

~

j:

11illill

0.01%

f=20Hz

j:

100

20

IIII

0.01%
0.01

10k 20k

1k
f - Frequency - Hz

0.1
Po - Output Power - W

FigureS

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

¥

t::

=:

f=

~

~

+

i

10%
RL=SQ
Ay = +20 toO dB
BTL

If

RL=SQ

t- Ay=+20toOdB

t- BTL

+

c

f0

%

~

!

I

~

Iz

.2

:c

F

,I

1%

19"

,
l"-

0

E
III

:c

1;1

Po=O.5W

""'"

f=20kHz

.~

Po = 0.25 W
0.1 %

10

~

0.1% I"'--

f= 1 kHz

I

~

~

j:
0.01'%
20

z+

r

Q

:c

I-

PO=1W
100

1k
f - Frequency - Hz

10k 20k

r-- f=20Hz
0.01%
0.01

1111111

Figure 7

0.1
Po - Output Power - W

FigureS

-!I
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

10

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL

SLOS286 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DlSTORnON PLUS NOISE

va

va

FREQUENCY

OUTPUT POWER
10%

RL=320
AV= +14 to 0 dB
SE

J0
z

+

1""

--

jfii .:;

.1
~

:;;;;;

Po·25mW

~

......

1

1%

=

'=20kHz

I
S
z

Jllill
100

j!:

1111111
1k
,- Frequency - Hz

r-- '=1kHz
r--...,
r-...

0

Po =75 mW

0.01%
0.01

10k 20k

-

0.1%

I

Po=50mW ~
0.00I'll
20

Ii

f=mHZ

0.1
Po - Output Power - W

Figure 9

Figure 10

TOTAL HARMONIC DISTORnON PLUS NOISE

TOTAL HARMONIC DISTORnON PLUS NOISE

va

vs

FREQUENCY

OUTPUT VOLTAGE

10%

10%

~ RL=10kO

f:

I

r-

+

II

·z1

AV=+14toOdB
SE

+

I

1%

If

~

1%

i

I

0.1%
VO=1 VRMS

I

RL=320
Av=+14toOdB
SE

ll.

0.1%

S

0.01%

t-

~

...

100

1k
,- Frequency - Hz

10k 20k

-- ..

-~

0.001%

o

,
I
1

...

RL=10kO
AV=+14toOdB
SE

j!:
0.001%
20

f=2OkHz

,=~ kH;-

0.01%

I

j!:

~ lS..

'=2OHz

1 J

0.2 OA 0.6 0.8
1 1.2 1A 1.6 1.6
Vo - Output Voltage - VRMS

Figure 11

Figure 12

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

2

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

160
1/1

140

~

120

vs

FREQUENCY

FREQUENCY

,

V~D~'5V

SUPPLY RIPPLE REJECTION RATIO

vs
0

'"

BW = 22 Hz to 22 kHz
RL=40

I

III

"

'S

60

~

/

;;
20

.....

o

o

ia:
c

ta:

"ii'

IIIIII

it

-

~
CI.

100

-40

~O

I'

.....

...",::

I\,

CI.

I'"
AV=+6dB

I

AV = +20 dB

t

"",,10-

40

I

1-""1-'

AV= +20 dB
60

0

V

100

!z

-20

I

I

s&
j

RL=SO
CB=OA7J.LF
BTL

-80

AV=+6dB

:::I

1/1

1k
f - Frequency - Hz

-100
-120

10k 20k

100

20

Figure 13

1k
f - Frequency - Hz

III

"

t
I:

FREQUENCY

'

,

-50
~O

1'""
-40

...........

c
.2

ia:

vs

FREQUENCY

CB=0.47J.LF
-20 I- SE

I

j

1'1'

~O

1/1

AV=+6dB

~

III

".oc

t)

-«I

P~~','W

,

RL=SO
AV= +20 dB
BTL

-70

L
LEFT TO RIGHT

/""

I

I

~ P"

-80

V

~j;'

--90

RIGHT TO LEFT

AV=+14dB
II

.2CI.
CI.
:::I

CROSSTALK

vs
-40

RL'~ 3'2'0

10k 20k

. Figure 14

SUPPLY RIPPLE REJECTION RATIO

0

Ir

-100

-100

-110

-120
20

100

1k
f - Frequency - Hz

10k 20k

-120
20

100

1k

10k 20k

f - Frequency - Hz

Figure 15

Figure 16

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-«i1

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
CROSSTALK

CROSSTALK

vs

vs

FREQUENCY

FREQUENCY

-40

0
PO=1W
RL=Sn
AV=+6dB
BTL

-50
-60
III

"I

-70

ie

-80

II ~ll

V

LEFT TO RIGHT

~

CJ

-20

-90

I I

r-

.....

JI1mtif

J...'i-"

./

VO=1 VRMS
RL=10kn
AV=+6dB
SE

-40

III

"I

~

~

-60

~

LEFT TO RIGHT

CJ

-80 1'00.

-100

I

-110
-120
20

100

1k
f - Frequency - Hz

-120 L....l....w...L.I.1.LL.---L....J....I..I...I..I..LI.L._J.....,J....w....I...U.Uo:--'
20
100
1k
10k 20k

10k 20k

f - Frequency - Hz

Figure 17

Figure 18

SHUTDOWN ATTENUATION

SIGNAL·TO-NOISE RATIO

vs

vs

FREQUENCY

FREQUENCY

0
VI=1 VRMS

ilII

-20
III

"cI

i

III

~

RL = 10 kn, SE

-40

C

11
::J

-60

-80

.c

",...

-100

~

I-

RL=Sn,B+L
-120
20

3-652

~

RL=32n,SE

II)

"I

J
.~
~

::J

~
~

~;~~~ T01L~~

-100

120 1"""T"T"TT1Trr-"""''''''"TTTTTT"-Tl"T"1"""1''TTrr-.
PO=1W
115 RL=Sn
BTL
110 H+t+ltIt--++1H+tttt-HH-1I+tttt--I

105

~

1
I

a:

z
II)

II

II I I"" I"1k

100

10k 20k

f - Frequency - Hz

f - Frequency - Hz

Figure 19

Figure 20

~TEXAS .
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER 1999

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE
30

1~~I~I~nl II

25 _

IIIII

AV=+20dB
BTL

ri~:~1

20
15

U~:

")1I

IIIII

I

c

.~

01

!\

Phase

I--.

90'

1\

:E
0°

~

r\

5

I

....E

~

o

-10
10

il

.c

-900

-180°
100

1k

10k

1M

100k

f - Frequency - Hz

Figure 21

CLOSED LOOP RESPONSE
30

'"'

RL=8n
AV=+6dB
BTL

25

90°

20

c

.~

15
ID

'D

I

.j

10

01

'roo

:E

Phase
'III

0°

I I II

c:J
5

~

0

....E
-90°

"

-5
-10
10

I

\~

Gain

!I

.c

~

_180°
100

1k

10k

100k

1M

f - Frequency - Hz

Figure 22

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-653

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286-NOVEMBER 1999

TYPICAL CHARACTERISTICS
OUTPUT POWER
VB
LOAD RESISTANCE

OUTPUT POWER
VB
LOAD RESISTANCE

3.5

1500

AV=+20toOdB
BTL
3

:=I

1

\

2

10%THD+N

~
\~
\.

1.5

'!i

1250 II

\

2.5

Ii

AV= +14 to 0 dB

0

I

~

I

750

I

IIIII

o

1000

t
0

1%THD+N

o

~I
'!i

I"- t-~

0.5

SE

8

16

~

250

"""

24

~
~

500

32

40

48

56

o

64

~ 10%THD+N
~

1%THD+N
I
-.l
8
16

o

RL - Load Resistance - 0

Figure 23

L~

c

0

I

1.4

I

1.2

-

j

I
c

0.8
0.6

D.

0.4

~

40

V~

0.35

:=I

---

c
.S!

I

80

V

I

--

~

Each Channel

1

1.5

2

I

0.25

66

64

r-.... ~O
~

V

/

0.2
0.15

.....

IL ~ ~

80

fl

.........

0.1

r"

320
0.05 ~
I'

f=1kHz
BTL
0.5

L

0.3

I

~

0.2

o
o

48

OA

IL

c
D.
I

30

lL::=

//V'

40

POWER DISSIPATION
VB
OUTPUT POWER

1.8

:=I

32

Figure 24

POWER DISSIPATION
VB
OUTPUT POWER

1.6

24

RL - Load Resistance - 0

2.5

o
o

f=1kHz
BTL
Each Channel

~

Po - Output Power - W

U

U

M

~

U

Po - Output Power - W

Figure 26

Figure 25

,~.lExAs

INSTRUMENTS

POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

M

U

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286-NOVEMBER 1999

TYPICAL CHARACTERISTICS
POWER DISSIPATION

VB

AMBIENT TEMPERATURE

GAIN

7
6

\

~

c

iis

I

5

4
3

c

........

9JA1,2

II.

I

""

jJA3,

2

II.

o

~~

0

~

i\

c::

...

70

fl
c

60

I.Ii

50

'SII.

@

01

1\

""'

~

80

I

\,
"~ t\.
~
........

@

-- "" '\

90

.!.

9JA1 = 45.9°CJW
9JA2 = 45.2°CJW
9JA3 = 31.2°CJW 9JA4 = 18.6°CJW

\

9JA4

I

INPUT IMPEDANCE

VB

M

r\

\

.5
I

N

~~

1001~m1~

\

30

\ '\

20
10
-40

-30

TA - Ambient Temperature - °C

-20

-10

o

10

~

AV-Galn-dB

Figure 27

Figure 28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-655

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286 - NOVEMBER 1999

THERMAL INFORMAnON
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 29)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them appliceble for surface-mount applicetions. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems,. and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame deSign and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Side View (a)

Thennal
Pad

EndVJew(b)

Bottom View (c)

Figure 29. Views of Thermally Enhanced PWP Package

3-656

:'I
TEXAS
INSTRUMENTS
POST OFFICE BOX 656303 • DAUAS. TEXAS 75265

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286 - NOVEMBER 1999

APPLICATION INFORMATION
Table 1. DC Volume Control
VOLUME (Terminal 3)
FROM

GAIN of AMPLIFIER
(dB)

(V)

TO
(V)

0
0.15

0.15
0.28

20
18

0.28

0.39

16

0.39

0.5

14

0.5

0.61

12

0.61

0.73

10

0.73

0.84

8

0.84
0.95

0.95
1.06

6
4

1.06

1.17

2

1.17

1.28

1.28

1.39

0
-2

1.39

1.5

-4

1.5

1.62

-6

1.62

1.73

-8

1.73

1.84

-10

1.84

1.95

-12

1.95

2.07

-14

2.07

2.18

-16

2.18

2.29

-18

2.29

2.41

-20

2.41

2.52

-22

2.52

2.63

-24

2.63

2.74

-26

2.74

2.86

-28

2.86

2.97

-30

2.97

3.08

-32

3.08

3.2

-34

3.2

3.31

-36

3.31

3.42

-38

3.42

3.54

-40

3.54

5

-85

selection of components
Figure 30 and Figure 31 are schematic diagrams of typical notebook computer application circuits.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-657

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL

SL0S286 - NOVEMBER 1999

APPLICATION INFORMATION
Right
Hea~

phone
Input
Signal

CIRHP
0.47J.1F

-1

R
23

RLiNEIN

MUX

ROUT+

8

21

RIN

CRIN
0.47J.1F

T
PC BEEP
14
Input
Signal CPCB
0.47J.1F

-=

---1

COUTR
330J.1F

PC-BEEP
ROUT-

PCBeep

16

VDD

VDD

100kQ

r~
-=-

VOLUME
CLK
SElBTL

CCLK
47nFT

Left CILHP
Head- 0.47 J.1F
phone
Input
Signal

-J

CILLINE

2

HPILINE

6

LHPIN

5

LLiNEIN

Galnl
MUX
Control

Depop
Circuitry
Power
Management

PVDD

18

VDD

19

BYPASS
SHUTDOWN

11

GND
L
MUX

22

-J

-:r

LOUT+

4

LOUT-

9

VDD
CSR
0.1 J.1F
VDD

T
-=

-:r

To
SystemControl

Left 0.47 J.1F
Line
Input
Signal

See Note A

1kO

-=

P

CSR
0.1J.1F

CBYP
0.47 J.1F
1 kQ

1,12,
13,24

-=

-=

COUTL
330J.1F

LIN
CLiN
0.47J.1F T

-=

100kQ
NOTE A.

A 0.1 J.1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals. a larger
electrolytic capacitor of 10 J.1F or greater should be placed near the audio power amplifier.

Figure 30. Typical TPA0232 Application Circuit Using Single-Ended Inputs and Input MUX

~TEXAS

3-658

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0232
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER 1999

APPLICATION INFORMATION
N/e
RI ht
g
Negative
Differential
Input
Signal

20

RHPIN

23

RLiNEIN

CIRIN0.4711F

---.'l
I

ROUT+

21

ROUT-

16

Right
CIRIN+
Positive 0.47 I1F
--.'l
8
RIN
Differential 71---"-1--'-':=...-----+.
Input
Signal
PC BEEP
14
Input
Signal CpCB

--1

PC-BEEP

l50kU

lcclK

3

VOLUME

17

ClK

15

sEiBTL

2

HPILINE

6

lHPIN

5

lLiNEIN

N/C

I~put

-1

r---~~~~~---;::~:=::=.J.,
Gain!
MUX
Control

CILIN
Left
Positive 0.47 11
10
Differential
Input
Signal

PVDD

18

See Note A

I--....!....!~+--'-"-----.-- VDD

Depop
Circuitry

CSR

'J'
0.1 I1F
VDD

VDD 19
Power
Management !-'B=:Y=P:,::'A=S=St-1'-'.1_--,
SHUTDOWN 22

r

CSR
O.lI1F
CBYP

'J' 0.47 I1F

To
SystemControl

CILlN0.47 I1F

~

1 kU

100 kU

-=- 47nFT
-=-

Signal

VDD

0.47 11F

VDD

left
Negative
Differential

PCBeep

1 kg

lOUT+

COUTl
330l1F

liN

lOUT-

9

100 kU
NOTE A.

A 0.1 I1F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 I1F or greater should be placed near the audio power amplifier.

Figure 31. Typical TPA0232 Application Circuit Using Differential Inputs

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

:Hl59

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER' 1999

APPLICATION INFORMATION

input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass filter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r------------

I
I
I

c
Input Signal

At

IN
RI
---1f--.....----='-'---1I-"IIV'v-*-I
R

Figure 32. Resistor on Input for Cut-Off Frequency

The input resistance at each gain setting is given in Figure 28.
The -3 dB frequency can be calculated using the following formula:

f

-

1

-3 dB - 2n C(R II RI)

(1)

If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addition, the order of the filter could be increased.

input capacitor, CI
In the typical application an input capacitor, CI, is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, C, and the input impedance of the amplifier, Z" form a
high-pass filter with the corner frequency determined in equation 2.

f

-

c(hlghpass) -

(2)

1

2nZ IN C,

~TEXAS
3-660

INSTRUMENTS

POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER 1999

APPLICATION INFORMATION

input capacitor, C, (continued)
The value of CI is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where ZI is 710 k.Q and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.
C _ _1_
I -

(3)

2nZ I fc

In this example, CI is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 !J.F. A further
consideration for this capacitor is the leakage path from the input source through the input network (CI) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. Note that it is important to confirm the capacitor polarity in the application.

power supply decoupling, Cs
The TPA0232 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 !J.F placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capaCitor of 10 !J.F or greater placed near
the audio power amplifier is recommended.

mid rail bypass capacitor, CBYP
The mid rail bypass capacitor, CBYP. is the most Critical capacitor and serves several important functions. During
startup or recovery from shutdown mode, CBYP determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the mid rail generation circuit internal to the amplifier, which appears as degraded PSRR and
THO+N.
Bypass capacitor, CBYP. values of 0.47!J.F to 1 !J.F ceramic or tantalum low-ESR capacitors are recommended
for the best THO and noise performance.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-661

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL

SL0S286-NOVEMBER 1999

APPLICATION INFORMATION

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cd is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

(4)

fc(hlgh)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 I!F is chosen and loads vary from 3 n,
4 n, 8 n, 32 n, 10 kO, and 47 kn Table 2 summarizes the frequency response characteristics of each
configuration.

Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode

Cc

Loweat Frequency

30

330!1F

161 Hz

40

330!1F

120Hz

SO

330!1F

60Hz

320

330!1F

15 Hz

10,0000

330!1F

0.05 Hz

47,0000

330!1F

0.Q1 Hz

RL

As Table 2 indicates, most of the bass response is attenuated into a 4-n load, an 8-n load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled Simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

~TEXAS

INSTRUMENTS
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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0232
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286-NOVEMBER 1999

APPLICATION INFORMATION

bridged-tied load versus single-ended mode
Figure 33 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0232 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but, initially consider power to the load. The differential drive to the speaker
means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the
voltage swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power
equation, where voltage is squared, yields 4x the output power from the same supply rail and load impedance
(see equation 5).

v

_ VO(PP)

(nns) -

(5)

212

2
V(nns)

Power = - RL

Voo

oJ' :

RL

Voo

J'!
rv :

VO(PP)

2x VO(PP)

-VO(PP)

Figure 33. Bridge-Tied Load Configuration

~1ExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

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TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL

SLOS286 - NOVEMBER 1999

APPLICATION INFORMATION
In a typical computer sound channel operating at 5 V, bridging raises the power into an a-n speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 34. A coupling capacitor is required to block the
dc offset voltage from reaching .the load. These capacitors can be quite large (approximately 3311F to 1000 11F)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
f

-

(c) -

1

(6)

2nRL C c

For example, a 6a-I1F capacitor with an a-n speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.

voo

~dB~-----J~=====

Figure 34. Single-Ended Configuration and Frequency Response

Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

slngle-ended operation
In SE mode (see Figure 33 and Figure 34), the load is driven from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SElBTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier'S gain by 6 dB.

Input MUX operation
The input MUX allows two separate inputs to be applied to the amplifier. This allows the designer to choose
which input is active independent of the state of the SElBTL terminal. When the HPILINE terminal is held high,
the headphone inputs are active. When the HP/LINE terminal is held low, the line BTL inputs are active.

~TEXAS

3-664

INSTRUMENTS
POST OFFICE BOX _ . DAllAS. TEXAS 75265

TPA0232
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286 - NOVEMBER 1999

APPLICATION INFORMATION

BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The internal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 35).
vo

'00

,/

,/
-~- 'OO(avg)

V(LRMS)

Figure 35. Voltage and Current Waveforms for BTL Amplifiers

Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a full-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
Efficiency of a BTL amplifier =

P
p-'=-

(7)

SUP

Where:
Vp
r;;'

,,2

and Psup = Voo looavg

and

Vp

looavg

f

=k

Therefore,
P
_ 2 Voo Vp
SUP It RL

Jt

o

2

~

therefore, PL =

L

V

RP sin(t) dt

L

=kx

substituting PL and Psup into equation 7,
Efficiency of a BTL amplifier

Where:

V p2
It Vp
2 RL
2Voo Vp = 4 Voo
It RL

V

:rr

RP[cOS(t)] 0
L

=

2Vp
:rr RL

PL = Power delivered to load
Psup =Power drawn from power supply
VLRMS =RMS voltage on BTL load
RL =Load resistance
Vp =Peak voltage on BTL load
looavg =Average current drawn from
the power supply
VOO =Power supply voltage
T]BTL =Efficiency of a BTL amplifier

Therefore,
TJBTL

(8)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

3-665

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286-NOVEMBER 1999

APPLICATION INFORMATION
Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the intemal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-0 loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 3. Efficiency Vs Output Power in S-V 8-0 BTL Systems
Output Power

t

Efficiency
(%)

Peak Voltage

(W)

(V)

Internal Dissipation

(W)

0.25

31.4

2.00

0.55

0.50
1.00

44.4
62.8

2.83
4.00

0.62
0.59

1.25

70.2

4.47t

0.53

High peak voHages cause Ihe THO 10 increase.

A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

crest factor and thermal considerations
Class-AB power amplifiers dissipate a significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of tM signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the intemal
dissipated power at the average output power level must be used. From the TPA0232 data sheet, one can see
that when the TPA0232 is operating from a 5-V supply into a 3-0 speaker that 4 W peaks are available.
Converting watts to dB:

P
PdB = 10Log ---.1Y. = 10Log 4 W = 6 dB
Pref
1W
Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6 dB 6 dB 6 dB 6 dB 6 dB -

15 dB = -9 dB (15 dB crest factor)
12 dB =-6 dB (12 dB crest factor)
9 dB =-3 dB (9 dB crest factor)
6 dB = 0 dB (6 dB crest factor)
3 dB = 3 dB (3 dB crest factor)

~TEXAS

3-666

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

(9)

TPA0232
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286-NOVEMBER 1999

APPLICATION INFORMATION
Converting dB back into watts:
P w = 10PdB/10 x P ref

(10)

= 63 mW (18 dB crest factor)
= 125 mW (15 dB crest factor)

=
=

250 mW (9 dB crest factor)
500 mW (6 dB crest factor)

= 1000 mW (3 dB crest factor)
= 2000 mW (15 dB crest factor)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. Using the power dissipation curves for as-V, 3-0 system, the internal dissipation in the TPA0232 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0232 Power Rating, 5-Y, 3-n, Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE
-3°C

4

2W(3dB)

1.7

4

1000 mW (6 dB)

1.6

6°C

4

500 mW (9 dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

78°C

4

63 mW(18 dB)

0.6

96°C

Table 5. TPA0232 Power Rating, SOY, 8-n, Stereo
PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

2.5W

1250 mW (3 dB crest factor)

0.55

100°C

2.5W

1000 mW (4 dB crest factor)

0.62

94°C

2.5W

500 mW (7 dB crest factor)

0.59

97°C

2.5W

250 mW (10 dB crest factor)

0.53

102°C

The maximum dissipated power, POmax, is reached at a much lower output power level for an 8 0 load than for
a 3 0 load. As a result, this simple formula for calculating POmax may be used for an 8 0 application:

= 2V50

P
Omax

(11)

:rt2RL

However, in the case of a 3 0 load, the POmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 0 load.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS. TEXAS 75265

3-667

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286-NOVEMBER 1999

APPLICATION INFORMATION
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to 0JA:
e

JA

=

1
Derating Factor

= _1_
0.022

= 450C/W

(12)

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated power needs to be doubled for two channel operation. Given 0JA, the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0232 is
150°C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
TA Max = T J Max - e JA Po

(13)

= 150 - 45(0.6 x 2) = 96°C (15 dB crest factor)
NOTE:

Internal diSSipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.
Tables 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0232 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Table 4 and 5 were calculated for maximum listening
volume without distortion. When the output level is reduced the numbers in the table change significantly. Also,
using
speakers dramatically increases the thermal performance by increasing amplifier efficiency.

8-n

SE/BTL operation
The ability of the TPA0232 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
intemal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0232, two separate amplifiers drive OUT+ and OUT-. The SEIBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When
SE/BTL is held low, the amplifier is on and the TPA0232 is in the BTL mode. When SE/BTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0232 as an SE driver from LOUT+
and ROUT+(terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SE/BTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 36.

~ThxAs

INSTRUMENTS
3-668

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S286-NOVEMBER 1999

APPLICATION INFORMATION

20

RHPIN

23

RLiNEIN
ROUT+

8

21

RIN

VDD
ROUT-

16

100kn
SE/Bl'[

15 100 kn

~

n

,----~
Figure 36. TPA0232 Resistor Divider Network Circuit

Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 100-kil/1-kQ divider pulls the SElBTL input low. When a plug is inserted, the 1-kO
resistor is disconnected and the SElBTL input is pulled high. When the input goes high, the OUT-amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (CO) into the headphone jack.

PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few extemal components. The input is normally activated automatically. When the PC BEEP input
is active, both of the LlNEIN and HPIN inputs are deselected and both the left and right channels are driven in
BTL mode with the signal from PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 VN
and is independent of the volume setting. When the PC BEEP input is deselected, the amplifier will return to
the previous operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating
PC BEEP will take the device out of shutdown and output the PC BEEP signal, then return the amplifier to
shutdown mode.
The preferred input signal is a square wave or pulse train with an amplitude of 1 V pp or greater. To be accurately
detected, the signal must have a minimum of 1 Vpp amplitude, rise and fall times of less than 0.1 JlS and a
minimum of 8 rising edges. When the signal is no longer detected, the amplifier will return to its previous
operating mode and volume setting.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

3-669

TPA0232
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS286-NOVEMBER 1999

APPLICATION INFORMATION
If it is desired to ac-couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
the following equation:
C

PCB

~

271:

1

(14)

f PCB (100 kO)

The PC BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no signal is present.

shutdown modes
The TPA0232 employs a shutdown mode of operation designed to reduce supply current, Ipp, to the absolute
minimum level during periods of nonuse for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, Ipp = 150 IJA. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.
Table 6. HPILINE, SElBTL, and Shutdown Functions
AMPUFIER STATE

INPUTSt
HPILINE

SElBTL

SHUTDOWN

INPUT

OUTPUT

X

X

Low

X

Mute

Low

Low

High

Line

BTL

Low

High

High

Line

SE

High

Low

High

HP

BTL

High

High

High

HP

SE

t Inpuls should never be left unconnected.

=

X do not care

3-670

:IlJ1ExAs
INSTRUMENTS
POST OFFICE BOX 865303 • DALLAS. TEXAS 75285

TPA0233
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS278A - JANUARY 2000 - REVISED MARCH 2000

• Ideal for Notebook Computers, PDAs, and
Other Small Portable Audio Devices
• 2 W Into 4-0 From S-Y Supply
• 0.6 W Into 4-0 From 3-Y Supply
• Stereo Head Phone Drive
• Mono (BTL) Signal Created by Summing
Left and Right Signals
• Wide Power Supply Compatibility
3YtoSY
·3YtoSY
• Meets PC99 Portable Specs (target)
• Low Supply Current
- 4 mA Typical at S Y
- 3.3 mA Typical at 3 Y
• Shutdown Control ••• 1 ItA Typical
• Shutdown Pin Is nL Compatible
• -40°C to 8SoC Operating Temperature
Range
• Space-Saving, Thermally-Enhanced MSOP
Packaging

DGQPACKAGE
(TOP VIEW)
FILT_CAP

SHUTDOWN

Voo
BYPASS

RIN

LOIMO
LIN
GND
SRIMN
ROIMO

description
The TPA0233 is a 2-W mono bridge-tied-Ioad (BTL) amplifier designed to drive speakers with as low as 4-0
impedance. The mono signal is created by summing left and right inputs. The amplifier can be reconfigured
on-the-fly to drive two stereo single-ended (SE) signals into head phones. This makes the device ideal for use
in small notebook computers, POAs, digital personal audio players, anyplace a mono speaker and stereo head
phones are required. From a 5-Y supply, the TPA0233 can delivery 2-W of power into a 4-0 speaker.
The gain of the input stage is set by the user-selected input resistor and a 50-kO internal feedback resistor
AFt AI)' The power stage is internally configured with a gain of -1.25 VN in SE mode, and -2.5 VN in
(Av
BTL mode. Thus, the overall gain of the amplifier is 62.5 kill AI in SE mode and 125 kill AI in BTL mode. The
input terminals are high-impedance CMOS inputs, and can be used as summing nodes.

=-

The TPA0233 is available in the 10-pin thermally-enhanced MSOP package (OGQ) and operates over an
ambient temperature range of -40°C to 85°C.

•

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

~lExAs

Copyright © 2000, Texas Instruments Incorporated

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-671

TPA0233
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SLOS278A - JANUARY 2000 - REVISED MARCH 2000

,-------------4

BVPASS--------l

3 1 VDD

VDD

I
I
I
I
I
I

rjFILTCAP
BYPASS

T-=- :I

SOkn
1.25*R

I
I

SI RIN
CI

Input

II--",RNIv-->

Audio
Inpm

CI

I
I
I
I
I
I

BYPASS

50kn
StereolMono
Control

50kn

STIMN I 7
I
I
I
I
I
I

1.2S*R

I

1~~RI~-9~1-L1-N~_1

From
System Control

Cc

ROIMO+ I 6

I
I
I
I
I
I
I
I
I
I
I
I
I
I

Left

VDD

1100kn

I
Right
Audio

8

GND

I
1I

-=-

Cc

LOIMO- 110

I
I
I
I
I

I
I
BYPASS

21 SHUTDOWN

Shutdown
andDapop
Circuitry

L _________________________

I
I
I
I
I
I
I
I

~

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

MSOpt
(DGQ)

-40"C to 85°C

TPA02330GQ

MSOP
SYMBOLIZATION
AEJ

t The OGQ package are available taped and reeled. To order a taped and reeled part, add the
suffix R to the part number (e.g., TPA0233DGQR).

~TEXAS

3-672

INSTRUMENTS
POST OFFICE sox 655303 • DALLAS. TEXAS 75265

1 kn

TPA0233
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S278A - JANUARY 2000 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME
NO.

DESCRIPTION

1f0

MONO·IN

1

I

SHUTDOWN

2

I

Mono input terminal
SHUTDOWN places the entire device in shutdown mode when held low. TTL compatible input.

VOO
BYPASS

3

I

VOO is the supply voltage terminal.

4

I

BYPASS is the tap to the voltage divider for intemal mid-supply bias. This terminal should be connected
to a O.I-J!F to l-J!F capacitor.
Right-channel input terminal

RIN

5

I

ROIMO

6

0

Right-output in SE mode and mono positive output in BTL mode

SRIMN

7

I

Selects between stereo and mono mode. When held high. the amplHier is in SE stereo mode. while held
low. the amplifier is in BTL mono mode.

GNO

8

LIN

9

I

Left-channel input terminal

LOIMO

10

0

Left-output in SE mode and mono negative output in BTL mode.

Ground terminal

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§
Supply voltage. Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are strass ratings only. and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device raliability.
DISSIPATION RATING TABLE
PACKAGE
OGO

DERATING FACTOR
2.14 w11

17.1 mW/"C

1.37W

1.11 W

'l1 Please see the Texas Instruments document. PowerPAD Thermally Enhanced Package Application Report
(literature number SLMA002). for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage. VOO
High-level input voltage, VIH

STIMN

IVOO=3V
I VOO=5V

SHUTDOWN
Low-level input voltage. VIL

STIMN

MIN

MAX

2.5

5.5

UNIT
V

2.7
V

4.5
2

IVOO=3V
IVOO=5V

1.65
2.75

SHUTDOWN
Operating free-air temperature, TA

V

0.8

-40

85

°C

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-673

TPA0233
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S278A - JANUARY 2000 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, VDD = 3 V, TA = 25°C (unless otherwise
noted)
.
TEST CONDITIONS

PARAMETER
IVool

Output offset voltage (measured differentially)

100

Supply current

IOD(SO)

Supply current, shutdown mode

operating characteristics, VDD

MIN

TYP

MAX

UNIT

30

mV

3.3

4.5

rnA

1

10

j.tA

TYP

MAX

=3 V, TA =25°C, RL =4 n

PARAMETER

TEST CONDITIONS
THD=1%,

BTL mode

THD=0.1%,

SEmode,

Po

Output power, see Note 1

THD+N

Total harmonic distortion plus noise

Po = 500 mW,

f= 20 Hz to 20 kHz

BOM

Maximum output power bandwidth

Gain =2,

THO=2%

MIN

660

mW

33

RL=32n

UNIT

0.3%
20

kHz

NOTE 1: Output power is measured at the output terminals of the device at f = 1 kHz.

electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

IVOol

Output offset voltage (measured differentially)

100

Supply current

IOO(SO)

Supply current, shutdown mode

operating characteristics, VDD

TYP

MAX

UNIT

30

mV

4

5

rnA

1

10

j.tA

=5 V, TA =25°C, RL =4 n

PARAMETER

TEST CONDITIONS
THD = 1%,

BTL mode

THD=0.1%,

SEmode,

Po

Output power, see Note 1

THO+N

Total harmonic distortion plus
noise

PO=1 W,

f=20 Hz to 20 kHz

BOM

Maximum output power bandwidth

Gain =2.5,

THD=2%

MIN

RL=32n

NOTE 1: Output power is measured at the output terminals of the device at f = 1 kHz.

~TEXAS

3-674

MIN

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

TYP

MAX

UNIT

2

W

92

mW

0.2%
20

kHz

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
• Compatible With PC 99 Desktop Line-Out
Into 10-kn Load
• Compatible With PC 99 Portable Into 8-n
Load
• Internal Gain Control, Which Eliminates
External Gain-Setting Resistors
• DC Volume Control From 20 dB to -40 dB
• 2-W/Ch Output Power Into 3-n Load
• Input MUX Select Terminal
• PC-Beep Input
• Depop Circuitry
• Stereo Input MUX
• Fully Differential Input
• Low Supply Current and Shutdown Current
• Surface-Mount Power Packaging
24-Pin TSSOP PowerPADTM

PWPPACKAGE
(TOP VIEW)

GND
HP/LINE
VOLUME
LOUT+
LLiNEIN
LHPIN
PVoo
RIN
LOUTLIN
BYPASS
GND

10
2
3
4
5
6
7
8
9
10
11
12

24
23
22
21
20
19
18
17
16
15
14
13

GND
RLiNEIN
SHUTDOWN
ROUT+
RHPIN
Voo
PVoo
CLK
ROUTSElBTL
PC-BEEP
GND

description
The TPA0242 is a stereo audio power amplifier in a 24-pin TSSOP thermally enhanced package capable of
delivering 2 W of continuous RMS power per channel into 3-n loads. This device minimizes the number of
external components needed, which simplifies the design and frees up board space for other features. When
driving 1 W into 8-n speakers, the TPA0242 has less than 0.22% THD+N across its specified frequency range.
Included within this device is integrated depop circuitry that virtually eliminates transients that cause noise in
the speakers.
Amplifier gain is controlled by a dc voltage input on the VOLUME terminal. There are 31 discrete steps covering
the range of 20 dB (maximum volume setting) to -40 dB (minimum volume setting) in 2 dB steps. When the
VOLUME terminal exceeds 3.54 V, the device is muted. An internal input MUX allows two sets of stereo inputs
to the amplifier. The HPILINE terminal allows the user to select which MUX input is active regardless of whether
the amplifier is in SE or BTL mode. In notebook applications, where internal speakers are driven as BTL and
the line outputs (often headphone drive) are required to be SE, the TPA0242 automatically switches into SE
mode when the SElBTL input is activated, and this effectively reduces the gain by 6 dB.
The TPA0242 consumes only 20 mA of supply current during normal operation. A miserly shutdown mode
reduces the supply current to less than 150 IlA.
The PowerPAD package (PWP) delivers a level of thermal performance that was previously achievable only
in TO-220-type packages. Thermal impedances of approximately 35°C/W are truly realized in multilayer PCB
applications. This allows the TPA0242 to operate at full power into 8-n loads at ambient temperatures of 85°C.

A.

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAO is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

Copyright © 1999, Texas Instruments Incorporated

3-675

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL

SL0S287-NOVEMBER 1999

functional block diagram

~

RHPIN

RLiNEIN - - - - - 1

M~X ""-....,........
>-""*-------

ROUT+

>-......+ - - - - - -

ROUT-

VOLUME - - - - - - -..

RIN --------+---+~

PC-BEEP -1L..-Beep_PC_...

ro:;:;;-,
SElBTL
HPIUNE

LHPIN

---j

---1

Power

~

Management

PVDD
VDD
BYPASS
SHUTDOWN

MUX
' - - - - - GND

Control

(;g--

LLiNEIN - - - - - 1 MtX

....._ - - 1

>--+--1------

LOUT+

>-......- - - - - -

LOUT-

LlN----------+~

~TEXAS

3-676

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

AVAILABLE OPTIONS
PACKAGED DEVICE
TA

TSSOpt
(PWP)

-40°C to 85°C

TPA0242PWP

t The PWP package IS available taped and reeled. To order a taped and reeled part,
add the suffix R to the part number (e.g., TPA0242PWPR).

Terminal Functions
TERMINAL
NAME

NO.

BYPASS

11

ClK

17

GND

1,12
13,24

110

DESCRIPTION
Tap to voltage divider for intemal mid-supply bias generator

I

If a 47-nF capacitor is attached, the TPA0242 generates an internal clock. An external clock can override
the intemal clock input to this terminal.
Ground connection for circuitry. Connected to thermal pad

lHPIN

6

I

Left channel headphone input, selected when SEIBTL is held high

LIN

10

I

Common left input for fully differential input. AC ground for single-ended inputs
Left channel line negative input, selected when SE/BTL is held low

LLiNEIN

5

I

LOUT+

4

LOUT-

9

0
0

HP/LINE

2

I

HPILINE is the input MUX control input. When the HP/LiNE terminal is held high, the headphone inputs
(LHPIN or RHPIN [6, 20)) are active. When the HPILINE terminal is held low, the line BTL inputs (LLINEIN
or RLiNEIN [5, 23)) are active.

PC-BEEP

14

I

The input for PC Beep mode. PC-BEEP is enabled when a > 1-V (peak-to-peak) square wave is input to
PC-BEEP.
Power supply for output stage

Left channel positive output in BTL mode and positive output in SE mode
Left channel negative output in BTL mode and high-impedance in SE mode

PVDD

7, 18

I

RHPIN

20

I

Right channel headphone input, selected when SElBTL is held high

RIN

8

I

Common right input for fully differential input. AC ground for single-ended inputs

RLiNEIN

23

I

Right channel line input, selected when SE/BTL is held low

ROUT+

21

ROUT-

16

0
0

Right channel negative output in BTL mode and high-impedance in SE mode

SElBTL

15

I

Hold SElBTL low for BTL mode and hold high for SE mode.

SHUTDOWN

22

I

When held low, this terminal places the entire device, except PC-BEEP detect circuitry, in shutdown mode.

VDD

19

I

Analog VDD input supply. This terminal needs to be isolated from PVDD to achieve highest performance.

I

VOLUME detects the dc level at the terminal and sets the gain for 31 discrete steps covering a range of
20 dB to -40 dB for dc levels of 0.15 V to 3.54. When the dc level is over 3.54 V, the device is muted.

VOLUME

3

Right channel positive output in BTL mode and positive output in SE mode

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAl.LAS, TEXAS 75265

3-677

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
Sl0S287 - NOVEMBER 1999

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
t Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions' is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
PWP

DERATING FACTOR
2.7wt

21.8mWf"C

1.7W

1.4W

:j: Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

(literature number SlMA002). for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage. VOO
High-level input voltage, VIH

MIN

MAX

4.5

5.5

SElBTl. HP/LINE

4

SHUTDOWN

2

SElBTl, HP/LINE

low-level input voltage, Vil

0.8

Operating free-air temperature, TA

-40

V
V

3

SHUTDOWN

UNIT

85

V
°C

=

electrical characteristics at specified free-air temperature, Voo = 5 V, TA 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS
VI = 0,

Supply ripple rejection ratio

VOO =4.9Vt05.1 V

IIIHI

High-level input current

Voo =5.5 V. VI=VOO

IIlll

low-level input current

VOO=5.5V. VI=OV

100

Supply current

IOO(SO)

Supply current, shutdown mode

TYP

20

SEmode

10
150

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

MAX

UNIT

25

mV

900

nA

900

nA

dB

67

BTL mode

~1ExAs

3-678

MIN

Av = 2 VN

Output offset voltage (measured differentially)

IVOSI

mA
300

ItA

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287-NOVEMBER 1999

operating characteristics, Voo = 5 V, TA = 25°C, RL = 4 n, Gain = 2 VN, BTL mode (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

Po

Output power

THO=I%,

f=lkHz

THO+N

Total harmonic distortion plus noise

PO=1 W,

f = 20 Hz to 15 kHz

BOM

Maximum output power bandwidth

THO=5%

Vn

MIN

TYP

MAX

UNIT

2

W

0.22%
>15

Supply ripple rejection ratio

f = 1 kHz, CB = 0.47 I1F

Noise output voltage

CB= 0.47 I1F,
f= 20 Hz to 20 kHz

BTL mode

65

SEmode

60

BTL mode

34

SEmode

44

kHz
dB

I1V RMS

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Output power
vs Voltage gain

1,4,6,8,10
2

THO+N

Total harmonic distortion plus noise

Vn

Output nOise voltage

VB Bandwidth

13

Supply ripple rejection ratio

VB Frequency

14,15

Crosstalk

VB Frequency

16,17,18

Shutdown attenuation

VB Frequency

19

SNR

Signal-to-noise ratio

VB Bandwidth

20

Po

Output power

VB Frequency
VB Output voltage

Closed loop response

Po

Power dissipation

ZI

Input impedance

3,5,7,9,11
12

21,22
vs Load resistance

23,24

VB Output power

25,26

vs Ambient temperature

27

VB Gain

28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAu.AS, TEXAS 75265

3-679

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287-NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

OUTPUT POWER

VOLTAGE GAIN

10"10

1%

.10

I

+

.L

z

c

I

0

'E

1%

i

i==

r-

.~

I

I
RL=4n!

t- Po = 1 W for Ay~B
~ yO= 1 YRMS for A~ dB
t- RL=8n

c

r-

~

E

I if

BTL

~

i.S!
=
c
0

I I

0.1%

iz

+

RL=3n

0

S

I

II

RL=8n

E
III

:I:

I

\

0.1%

III

:I:

'"........

S

~

~

I

....::

I-

0.01%
0.5 0.75

1

1.25 1.5 1.75

Ay = +20 to 4 dB
f=1kHz
BTL
2

Z

-

2.25 2.5 2.75

f'...

.:!i:I:
3

0.01%
-40

-30

Figure 1

-20
-10
o
Ay - Yoltage Gain - dB

10

20

Figure 2

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

OUTPUT POWER

10"10

10%

RL=3n
Ay= +20 to 0 dB
BTL

j

~

I-

Po - Output Power - W

RL=sn
Ay = +20 to +4 dB
BTL

GI

.s0
z

+

+

c

l5

!..
J
~

~

I

Z

.:!i:I:

r--,.....

.2

1::

1%
PO=0.5W ~
V .....

.~

"'"

0

PO=1W

E
:!

lt~

0.1 '!/

i

1%

~

If

I

0.1%

f= 20 kHz

I
..... fI!I=

E E

1

f= 1 kHz

f=~

100.

I"

V

z

~

PO=1.75W -

j!:

0.01 %
20

II jllli
100

1k
f - Frequency - Hz

.:!i

j!:

10k 20k

0.01%
0.01

Figure 3

0.1
Po - Output Power - W

Figure 4

~1ExAs

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

10

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE
VB
FREQUENCY

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
OUTPUT POWER

10%

10%

J0
z

+

c

+

c

0

1:

i
is

0

1:

1%

1%

ic

.!!
c
0

!III

:I:

!z

RL=40
Ay = +20 to +4 dB
BTL

·z1

RL=40
Ay = +20 to +4 dB
BTL

Po=O.25W

0.1%

f= 20 kHz

,

"0

~

C

V

I§

.... ~~

III

~

:I:

PO=1.5W -

Z

S

r - l - f-

I

+

c

r

i!:
0.01%
20

rliUill

I """

IIIIIII

1k
I - Frequency - Hz

1=20Hz

..
+

C
:I:

I

100

0.01%
0.01

10k 20k

0.1
Po - Output Power - W

FigureS

=
,.-

-

+

RL=80
Ay = +20 to +4 dB
BTL

i

t- BTL

j
I

c

.S!
PO=0.25W

i

t!i

RL=80

t- Ay = +20 to +4 dB

+

~

:I:

I

1=
10%~~EE.

~

1%

~0

!z

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
OUTPUT POWER

.~

c

~

10

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE
VB
FREQUENCY
10%

.!z

N

1=1 kHz
~

0.1%

~

I

L

~

0.1%

t::
~

1==

PO=0.5W

....
0.01%
20

=

~

i!:

PO=1W

100

0.1%~¥~~~~f=~1~k~H~zll~~11
E

'Ij

III"'"

i!:

1'0-

:I:

1k
I - Frequency - Hz

10k 20k

1= 20 Hz

I IIIIII

J--II'..--""t-~'H-!.III--+-H+tI1+tI

~

0.01% L--J.......I....u.II.l..LJ.II.LI.-....II--J...J,.................---'-..............
0.01
0.1
10
Po - Output Power - W

Figure 7

Figure 8

~TEXAS

INSTRUMENTS
POST OFFICE BOX 6!i5303 • DALlAS" TEXAS 75265

3-681

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTOFlnON PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

va

va

FREQUENCY

OUTPUT POWER

10%r:;:~em~
=

10%

RL=320
Ay +1410 +4 dB
SE

RL=320
Ay= +14 to +4 dB
SE

=

f=2OkHz

r--~

~=1kHz

r
0.01%
0.01

10k 20k

fll=1:::

f=20Hz
0.1
Po - Output Power - W

f - Frequency - Hz

Figure 9

Figure 10

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

va
10%

t=

RL=10kn
Ay=+14toOdB
r- SE

t=

I

I
+

.S!

g

1%

1%

~

I

YO=1 YRMS

I

OUTPUT VOLTAGE

+

I
z

j
I

0.1%

If

va

10%

FREQUENCY

0.01%

"'"

iJ

o
j!:
0.001%
20

r-- t-

r-

1k

10k 20k

f=1lkHz I
-.A

0.01%
RL=10kn
Ay +14 to +4 dB
SE

=

j!:
100

f=2OkHz

~

0.1%

0.001%
0.2 0.4

f - Frequency - Hz

0.6

0.8

f= 20 Hz

I I
1

1.2 1.4

1.6

YO - Output Yoltage - YRMS

Figure 11

Figure 12

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

1.8

2

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
OUTPUT NOISE VOLTAGE

SUPPLY RIPPLE REJECTION RATIO

vs

vs

BANDWIDTH

FREQUENCY
0

I

~

VOO=5V
RL=4n

140

IX!
'a

120 t-+-+t-ttttt--+-+-IH+tttt-+-It-Hf-t+ttt-----I

I

t
~
•
~

/

~~H*m-~~~-+~~~

II:

.S!

t>
Gi'
'"

a.
'CI."

H-++++t+1-

III I

-40

A7 +20 dB

c

II:

~

-20

I
0

i

100 t-+-++++H+--I-+++++l-H--l--+H-++H+---l

RL=8Q
CB =0.47 I1F
BTL

ii:

--60

r'\\
r--.,

V

-80

\

~
CI.
CI.
::I

III -100

o

100

1k
BW - Bandwidth - Hz

Illrii"l

-120

10k 20k

20

100

1k
f - Frequency - Hz

Figure 14

Figure 13
SUPPLY RIPPLE REJECTION RATIO

0

IX!
'a

vs

FREQUENCY

FREQUENCY

I

-eo

.........

I

r--.r-o

AVrOdB

Gi'-eo

II:

V

" ~~

.!I

1:
II:

IX!

-eo

i

'a

...

~

-80

S

-90

.

AV=+14dB

./

III I

-70

I

I

RL=8n
AV=+20dB
BTL

-50

r--r-o
-40

~~ ~111 V.J

I

CB=0.47 I1F
-20 I- SE

I
0

iII:

CROSSTALK

vs
-40

I~LI~~~ln

10k 20k

I--~
i'"~

LEFT TO RIGHT

RIGHT TO LEFT

-100

f""" V'

r-- I-Y

::I

fII -100

-110

-120

20

100

1k
f - Fraquency - Hz

10k 20k

-120
20

1k

100

10k 20k

f - Frequency - Hz

Figure 16

Figure 15

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-683

TPA0242
STEREO 2·WAUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
CROSSTALK

CROSSTALK

va

va

FREQUENCY

FREQUENCY

--40

-60
III

"I

1
S

--40

PO=1W
RL=8n
Ay=+6dB
BTL

-50

YO=1 YRMS
RL=10kn
Ay=+6dB
SE

-50

-60

111111

-70

I

-60

"

-70

I

.II<

I

II

11111
-90 I-

III

m~~TOIRIGHT

(.)

RIGHT TO LEFT

-90
-100

-110

-110
100

1k
f - Frequency - Hz

~

-120
20

10k 20k

"""

1k

Figure 18
SIGNAL-TO-NOISE RATIO

va

va

FREQUENCY

BANDWIDTH

0

120
Yl

z

1 YRMS

II

-20

III

"

I

I--'

.§

Z

-60

c

~
11:::I

"0I

~

--40

i:::I

!

RL=32o,SE
-60

1',....

ii
-100

J,...-

-

~

"-

105

r- r--.

I....... '"

~
c

~

95

II:

90

.......

i;

Ay = +20 dB

r-- I-t-

r--.,....

"""

Ay=+6 dB

I

Z

r---...

III

85

I 11111111
100

110

100

R L=8O, BTL

-120
20

PO=1W
RL=8n
BTL

115
RL=10kn,SE

III
0

10k 20k

f - Frequency - Hz

SHUTDOWN ATTENUATION

C

l"-

RIGHT TO LEFT

100

Figure 17

"

,

I Lilli

1"

-100

-120
20

LEFT TO RIGHT
-1

-60

1k
f - Frequency - Hz

10k 20k

80

o

Figure 19

100

1k
BW - Bandwidth - Hz

Figure 20

-!/}TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

10k 20k

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287-NOVEMBER 1999

TYPICAL CHARACTERISTICS

CLOSED LOOP RESPONSE

30

l~t~I~OI

25 r- Ay=+20dB

~~l~"

BTL

20

III
I

~

90°

II

~
15

If

'Q

180°

II

III

~

~~~:

......

10

.S
:IE

0°

r-...

J

11.
I

5

....E

o

-900

-s
-10
10

100

1k

10k

100k

1M

-180°

f - Frequency - Hz

Figure 21

CLOSED LOOP RESPONSE
30

180°

"""
RL=80
1111111

Ay=+6dB
BTL

25

I

~~~:

20

III

15

'Q

I

~

90°

I'-

i"I

V

10

......

r-...~

Gain

5

o

-s
-10
10

100

1k

10k

100k

1M

-180°

f - Frequency - Hz

Figure 22

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-685

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL

SL0S287 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
OUTPUT POWER
vs
LOAD RESISTANCE

OUTPUT POWER
vs
LOAD RESISTANCE

3.5
3

;=

1500

Ay= +20 to OdB
BTL

\

2.5

J \ 1\

~I

I

2

'5

t
0

1.5

10%THD+N

A-

'5
Go
'5

\~

I

0

~

rP
o

I

r'-. ~

0.5

1%THD+N

1000

7SO

~

SOO

~

rP
~~

250

" a I 16I I 24

32
40
46
RL - Load Resistance - n

o

,

--'

1250

I

56

o

64

AV= +14 to 0 dB
SE

~

10%THD+N

1%TH~
o

I

I

8

16
24
32
40
46
RL - Load Resistance - n

Figure 23

POWER DISSIPATION
vs
OUTPUT POWER

1.8

0.4

,~

1.8

I

c:
0

I
~

I

AI
Q
A-

1.4

I
//

1.2

0.8
0.8

OA

3n
0.35

/'
."...-

/1

V."

an

V

-

4n

o
o

;=

---

I

I
I
I
AI

-.

,p

0.5

1.5

2

Po - Output Power -

0.3
0.25
0.2
0.15
0.1

I

V

.....

/

1 I--

'L
'I

r---. r--!n
\""..

""-

,...8n

32n
0.05 ~['

f=1 kHz
BTL
Each Channel

0.2

2.5

w

o

o

~

" f'
f= 1 kHz
BTL
Each Channel

u

u

u

u

Po - Output Power -

Figure 25

Figure 26

~TEXAS

Ha6

64

Figure 24

POWER DISSIPATION
vs
OUTPUT POWER

;=

56

INSTRUMENTS
POST OFRCE BOX 655303 • DAlLAS, TEXAS 75.265

~

w

U

M

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287 - NOVEMBER 1999

TYPICAL CHARACTERISTICS
POWER DISSIPATION

vs
AMBIENT TEMPERATURE
7

I

\

9JA4

6

==I

r\

5

c

II

4

a.
I

jJA3,

"- ~

8JA1,2

.........

3
2

C

a.

o

~~

~

I

8JA1
8JA2
8JA3
8JA4

.1

.1

=45.9°CJW
=45.2°CJW
=31.2°CJW
=18.6°CJW

_

\
r\

""

1\,

~~
\
.........

~~

0 ~ ~ 00 00 ro01~1~100
TA - Ambient Temperature - °C

Figure 27
INPUT IMPEDANCE

vs
GAIN

90

80

c:

...

70

fl

60

I

c

-- ""

1\1

I.5
'!i

fI

~

N

30

"

\

50

\

\

\

20
10
~

-30

-20
-10
Av-Galn-dB

o

10

"

20

Figure 28

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

:Hl87

TPA0242·
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

THERMAL INFORMATION
The thermally enhanced PWP package is based on the 24-pin TSSOP, but includes a thermal pad (see Figure 29)
to provide an effective thermal contact between the IC and the PWB.
Traditionally, surface mount and power have been mutually exclusive terms. A variety of scaled-down TO-220-type
packages have leads formed as gull wings to make them applicable for surface-mount applications. These packages,
however, have only two shortcomings: they do not address the very low profile requirements «2 mm) of many of
today's advanced systems, and they do not offer a terminal-count high enough to accommodate increasing
integration. On the other hand, traditional low-power surface-mount packages require power-dissipation derating that
severely limits the usable range of many high-performance analog circuits.
The PowerPAD package (thermally enhanced TSSOP) combines fine-pitch surface-mount technology with thermal
performance comparable to much larger power packages.
The PowerPAD package is designed to optimize the heat transfer to the PWB. Because of the very small size and
limited mass of a TSSOP package, thermal enhancement is achieved by improving the thermal conduction paths that
remove heat from the component. The thermal pad is formed using a patented lead-frame design and manufacturing
technique to provide a direct connection to the heat-generating IC. When this pad is soldered or otherwise thermally
coupled to an external heat dissipator, high power dissipation in the ultra-thin, fine-pitch, surface-mount package can
be reliably achieved.

Side View (a)

Thermal
Pad

End View (b)

Bottom View (c)

Figure 29. Views of Thermally Enhanced PWP Package

3--688

-!11
TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

APPLICATION INFORMATION
Table 1. DC Volume Control
VOLUME (Terminal 3)
FROM
(V)

TO
(V)

GAIN of AMPLIFIER
(dB)

0

0.15

20

0.15

0.28

18

0.28

0.39

16

0.39

0.5

14

0.5

0.61

12

0.61

0.73

10

0.73

0.84

8

0.84

0.95

6

0.95

1.06

4

1.06

1.17

2

1.17

1.28

0

1.28

1.39

-2

1.39

1.5

-4

1.5

1.62

-6

1.62

1.73

-8

1.73

1.84

-10

1.84

1.95

-12

1.95

2.07

-14

2.07

2.18

-16

2.18

2.29

-18

2.29

2.41

-20

2.41

2.52

-22

2.52

2.63

-24

2.63

2.74

-26

2.74

2.86

-28

2.86

2.97

-30

2.97

3.08

-32

3.08

3.2

-34

3.2

3.31

-36

3.31

3.42

-38

3.42

3.54

-40

3.54

5

-85

selection of components
Figure 30 and Figure 31 are schematic diagrams of typical notebook computer application circuits.

!i1TEXAS

INSTRUMENTS

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-689

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287 - NOVEMBER 1999

APPLICATION INFORMATION
Right CIRHP
Head- 0.47 J!F
phone
Input
Signal

-1

8

ROUT+

21

ROUT-

16

RIN

CRIN
0.47J!F

T
-=PC BEEP
14
Input
Signal CPCB
0.47J!F
VDD

---j

i~~
-=-

CCLK
47nFT

PC-BEEP
PCBeep

100kn
VOLUME
CLK
SElBTL

Gain{
MUX

Control

2 HPILINE
Left CILHP ....,=-f-;:..:;;...:.='-"~--'
Head- 0.47 J!F
phone -11--+---,6=-f-L",H.!!.P-"IN-=-----t
Input
Signal
CILLINE ,-:5=-+-=="-1
Left 0.47J!F

Depop
Circuitry
Po_r
Management

PVDD

18

VDD

19

BYPASS
SHUTDOWN

11
22

See Note A

-:;r

VDD
CSR
0.1J!F
VDD

T

CSR
0.1J!F

CBYP
-:;r 0.47J!F

To
SyatemControl

1 kn

112
LOUT+

LIne -1
Input
Signal

-=-

COUTL
330J!F

LIN

LOUT-

9

100kn
NOTE A.

A 0.1 J!F ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 J!F or greater should be placed near the audio power amplifier.

Figure 30. Typical TPA0242 Application Circuit USing Single-Ended Inputs and Input MUX

~TEXAS

:HI90

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

APPLICATION INFORMATION
HI
20
Right
CIRINF
7
N
tl
0.4 1l
ega va ----'l
23
Differential I
Input
Signal

Right
PosHlve
Differential
Input
Signal

RHPIN
RLiNEIN
ROUT+

RIN

COUTR
330IlF

PC BEEP
Input ~ 1---'1",,4-+-,-==""""'''''-1
Signal CpCB

ROUT-

16

PVDD

18

lSOkn

lcclK
47nFT

3

VOLUME

17

ClK

lS

SEJiiT[

2

HPILINE

6

lHPIN

§]

r--_~~I/\A"'-'-+---;;~==-L.
Galnl
MUX
'--CO_ntr-ro_l....

Depop
Circuitry

Power
Management

NlC

VDD

19

BYPASS
SHUTDOWN

11

GND

left
Negative
IN.!-f
Differential -11-+-,STl",U",N",E:!!
I?put
Signal

VDD

1 kn

0.47 1lF

VDD

-=-

21

See Note A
VDD
CSR

-:J' O.lIlF

VDD

T

22

CSR
O.lIlF

CBYP

-:J' 0.471lF

To
SystemControl

1 kn

1,12,

CILlN0.471lF

COUTl
330IlF

Left
CILIN
PosHlve 0.47 11
Differential ~1-f-'1",-0+-l"'I"'N_ _ _ _
Input
Signal

-+*

lOUT-

9

lookn
NOTE A.

A 0.1 IlF ceramic capacitor should be placed as close as possible to the IC. For filtering lower-frequency noise signals, a larger
electrolytic capacitor of 10 IlF or greater should be placed near the audio power amplifier.

Figure 31. Typical TPA0242 Application Circuit Using Differential Inputs

~lExAs

INSTRUMENTS
POST OFFICE BOX 655303 • OALLAS, TEXAS 75265

3-691

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL~7-NOVEMBER100s

APPLICATION INFORMATION

Input resistance
Each gain setting is achieved by varying the input resistance of the amplifier, which can range from its smallest
value to over 6 times that value. As a result, if a single capacitor is used in the input high pass fiiter, the -3 dB
or cut-off frequency will also change by over 6 times. If an additional resistor is connected from the input pin
of the amplifier to ground, as shown in the figure below, the variation of the cut-off frequency will be much
reduced.

r------------

I
I
I

At

Input Signal ---1r-------'::.:-I---'w..~-I

R

Figure 32. Resistor on Input for Cut-Off Frequency
The input resistance at each gain setting is given in Figure 28:
The -3 dB frequency can be calculated using the following formula:

f

-3 dB - 211:

1

c( R " RI)

(1)

If the filter must be more accurate, the value of the capacitor should be increased while the value of the resistor
to ground should be decreased. In addltion, the order of the filter could be increased.

input capacitor, CI
In the typical application an input capacitor, C" is required to allow the amplifier to bias the input signal to the
proper dc level for optimum operation. In this case, C, and the input impedance of the amplifier, Z" form a
high-pass filter with the comer frequency determined in equation 2.

fC(highpass) =

(2)

23tZ~NCI

~1ExAs

3-692

INSTRUMENTS
POST OFFICE BOX 855303 • DALLAS. TEXAS 75285

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287 - NOVEMBER 1999

APPLICATION INFORMATION
The value of C, is important to consider as it directly affects the bass (low frequency) performance of the circuit.
Consider the example where Z, is 710 kn and the specification calls for a flat bass response down to 40 Hz.
Equation 2 is reconfigured as equation 3.

C I -

1

21tZ, fc

(3)

In this example, C, is 5.6 nF so one would likely choose a value in the range of 5.6 nF to 1 J.1F. A further
consideration for this capacitor is the leakage path from the input source through the input network (C,) and the
feedback network to the load. This leakage current creates a dc offset voltage at the input to the amplifier that
reduces useful headroom, especially in high gain applications. For this reason a low-leakage tantalum or
ceramic capacitor is the best choice. When polarized capacitors are used, the positive side of the capacitor
should face the amplifier input in most applications as the dc level there is held at Vool2, which is likely higher
than the source dc level. Note that it is important to confirm the capaCitor polarity in the application.

power supply decoupling, Cs
The TPA0242 is a high-performance CMOS audio amplifier that requires adequate power supply decoupling
to ensure the output total harmonic distortion (THO) is as low as possible. Power supply decoupling also
prevents oscillations for long lead lengths between the amplifier and the speaker. The optimum decoupling is
achieved by using two capacitors of different types that target different types of noise on the power supply leads.
For higher frequency transients, spikes, or digital hash on the line, a good low equivalent-series-resistance
(ESR) ceramic capacitor, typically 0.1 J.1F placed as close as possible to the device Voo lead, works best. For
filtering lower-frequency noise signals, a larger aluminum electrolytic capacitor of 10 J.1F or greater placed near
the audio power amplifier is recommended.

midrail bypass capacitor, CBYP
The midrail bypass capacitor, CSyp, is the most critical capacitor and serves several important functions. Ouring
startup or recovery from shutdown mode, CSyp determines the rate at which the amplifier starts up. The second
function is to reduce noise produced by the power supply caused by coupling into the output drive signal. This
noise is from the mid rail generation circuit internal to the amplifier, which appears as degraded PSRR and
THO+N.
Bypass capacitor, CSyp, values of 0.47 J.1F to 1 J.1F ceramic or tantalum low-ESR capaCitors are recommended
for the best THO and noise performance.

~TEXAS

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POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

3-693

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

APPLICATION INFORMATION

output coupling capacitor, Cc
In the typical single-supply SE configuration, an output coupling capacitor (Cc) is required to block the dc bias
at the output of the amplifier thus preventing dc currents in the load. As with the input coupling capacitor, the
output coupling capacitor and impedance of the load form a high-pass filter governed by equation 4.

(4)

fc(high)

The main disadvantage, from a performance standpoint, is the load impedances are typically small, which drives
the low-frequency corner higher, degrading the bass response. Large values of Cc are required to pass low
frequencies into the load. Consider the example where a Cc of 330 j.lF is chosen and loads vary from 3 n,
4
8 n, 32
10 kn, and 47 kn. Table 2 summarizes the frequency response characteristics of each
configuration.

n.

n.

Table 2. Common Load Impedances Vs Low Frequency Output Characteristics in SE Mode

Cc

Lowest Frequency

30

330l1F

161 Hz

40

33Ol1F

120Hz

80

33Ol1F

60Hz

320

33011F

15 Hz

10,0000

33O l1F

0.05 Hz

47,0000

330l1F

0.01 Hz

RL

4-n

8-n

As Table 2 indicates, most of the bass response is attenuated into a
load, an
load is adequate,
headphone response is good, and drive into line level inputs (a home stereo for example) is exceptional.

using low-ESR capacitors
Low-ESR capacitors are recommended throughout this applications section. A real (as opposed to ideal)
capacitor can be modeled simply as a resistor in series with an ideal capacitor. The voltage drop across this
resistor minimizes the beneficial effects of the capacitor in the circuit. The lower the equivalent value of this
resistance the more the real capacitor behaves like an ideal capacitor.

-!I
TEXAS
INSTRUMENTS
3-694

POST OFFICE BOX 655303 • DALLAS. lEXAS 75265

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

APPLICATION INFORMATION

bridged-tied load versus single-ended mode
Figure 33 shows a Class-AB audio power amplifier (APA) in a BTL configuration. The TPA0242 BTL amplifier
consists of two Class-AB amplifiers driving both ends of the load. There are several potential benefits to this
differential drive configuration, but, initially consider power to the load. The differential drive to the speaker
means that as one side is slewing up, the other side is slewing down, and vice versa. This in effect doubles the
voltage swing on the load as compared to a ground referenced load. Plugging 2 x VO(PP) into the power
equation, where voltage is squared, yields 4x the output power from the same supply rail and load impedance
(see equation 5).
VO(PP)

V(rms) =

(5)

2.f2
2

Power -

V(rms)

-~

Voo

J' :

RL

J'!

VO(PP)

2x VO(PP)

Figure 33. Bridge-Tied Load Configuration
In a typical computer sound channel operating at 5 V, bridging raises the power into an S-Q speaker from a
singled-ended (SE, ground reference) limit of 250 mW to 1 W. In sound power that is a 6-dB improvement which is loudness that can be heard. In addition to increased power there are frequency response concerns.
Consider the single-supply SE configuration shown in Figure 34. A coupling capacitor is required to block the
dc offset voltage from reaching the load. These capacitors can be quite large (approximately 33 JlF to 1000 JlF)
so they tend to be expensive, heavy, occupy valuable PCB area, and have the additional drawback of limiting
low-frequency performance of the system. This frequency limiting effect is due to the high pass filter network
created with the speaker impedance and the coupling capacitance and is calculated with equation 6.
(6)

~TEXAS

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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-695

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL

SL0S287 - NOVEMBER 1999

APPLICATION INFORMATION
For example, a 68-IlF capacitor with an 8-0 speaker would attenuate low frequencies below 293 Hz. The BTL
configuration cancels the dc offsets, which eliminates the need for the blocking capacitors. Low-frequency
performance is then limited only by the input network and speaker response. Cost and PCB space are also
minimized by eliminating the bulky coupling capacitor.
VDD

~dB~-----J~=====

Figure 34. Single-Ended Configuration and Frequency Response

Increasing power to the load does carry a penalty of increased internal power dissipation. The increased
dissipation is understandable considering that the BTL configuration produces 4x the output power of the SE
configuration. Internal dissipation versus output power is discussed further in the crest factor and thermal
considerations section.

single-ended operation
In SE mode (see Figure 33 and Figure 34), the load is driv~n from the primary amplifier output for each channel
(OUT+, terminals 21 and 4).
The amplifier switches single-ended operation when the SElBTL terminal is held high. This puts the negative
outputs in a high-impedance state, and reduces the amplifier's gain by 6 dB.

input MUX operation
The input MUX allows two separate inputs to be applied to the amplifier. This allows the designer to choose
which input is active independent of the state of the SElBTL terminal. When the HPILINE terminal is held high,
the headphone inputs are active. When the HP/LINE terminal is held low, the line BTL inputs are active.
BTL amplifier efficiency
Class-AB amplifiers are notoriously inefficient. The primary cause of these inefficiencies is voltage drop across
the output stage transistors. There are two components of the internal voltage drop. One is the headroom or
dc voltage drop that varies inversely to output power. The second component is due to the sinewave nature of
the output. The total voltage drop can be calculated by subtracting the RMS value of the output voltage from
Voo. The internal voltage drop multiplied by the RMS value ofthe supply current, loorms, determines the internal
power dissipation of the amplifier.
An easy-to-use equation to calculate efficiency starts out as being equal to the ratio of power from the power
supply to the power delivered to the load. To accurately calculate the RMS and average values of power in the
load and in the amplifier, the current and voltage waveform shapes must first be understood (see Figure 35).

~I
t TEXAS
3--696

NSTRUMENTS

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287 - NOVEMBER 1999

APPLICATION INFORMATION
'DO

,/
-~- 'OO(avg)

V(LRMS)

Figure 35. Voltage and Current Waveforms for BTL Amplifiers

Although the voltages and currents for SE and BTL are sinusoidal in the load, currents from the supply are very
different between SE and BTL configurations. In an SE application the current waveform is a half-wave rectified
shape, whereas in BTL it is a fUll-wave rectified waveform. This means RMS conversion factors are different.
Keep in mind that for most of the waveform both the push and pull transistors are not on at the same time, which
supports the fact that each amplifier in the BTL device only draws current from the supply for half the waveform.
The following equations are the basis for calculating amplifier efficiency.
P

Efficiency of a BTL amplifier = ~

(7)

SUP

Therefore,
_ 2 Vee Vp
P SUP -

It

RL

substituting PL and PSUP into equation 7,
V

Efficiency of a BTL amplifier =
Where:

P

2

PL = Power delivered to load
Psup = Power drawn from power supply
VLRMS =RMS voltage on BTL load
RL =Load resistance
Vp =Peak voltage on BTL load
leeavg =Average current drawn from
the power supply
Vee = Power supply voltage
l1BTL =Efficiency of a BTL amplifier

2Fil:

2 Vee Vp
It RL

Therefore,

(8)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-697

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLO~7-NOVEMBER1~

APPLICATION INFORMATION
Table 3 employs equation 8 to calculate efficiencies for four different output power levels. Note that the efficiency
of the amplifier is quite low for lower power levels and rises sharply as power to the load is increased resulting
in a nearly flat internal power dissipation over the normal operating range. Note that the internal dissipation at
full output power is less than in the half power range. Calculating the efficiency for a specific system is the key
to proper power supply design. For a stereo 1-W audio system with 8-n loads and a 5-V supply, the maximum
draw on the power supply is almost 3.25 W.
Table 3. Efficiency Vs Output Power In S-V 8-n BTL Systems
Output Power
(W)

Efficiency
(%)

Peek Voltage
(V)

0.25

31.4

2.00

0.55

0.50

44.4

2.83

0.62

1.00

62.8

4.00

0.59

1.25

70.2

4.47t

0.53

Intemal DIssipation
(W)

t High peak voltages cause the THO to Increase.

A final point to remember about Class-AB amplifiers (either SE or BTL) is how to manipulate the terms in the
efficiency equation to utmost advantage when possible. Note that in equation 8, Voo is in the denominator. This
indicates that as Voo goes down, efficiency goes up.

crest factor and thermal considerations
Class-AB power amplifiers dissipate a Significant amount of heat in the package under normal operating
conditions. A typical music CD requires 12 dB to 15 dB of dynamic range, or headroom above the average power
output, to pass the loudest portions of the signal without distortion. In other words, music typically has a crest
factor between 12 dB and 15 dB. When determining the optimal ambient operating temperature, the internal
dissipated power at the average output power level must be used. From the TPA0242 data sheet, one can see
that when the TPA0242 is operating from a 5-V supply into a 3-n speaker that 4 W peaks are available.
Converting watts to dB:
PdB

P

= 10Log~ =
Pref

10Log 4
1 Ww

=

6 dB

(9)

Subtracting the headroom restriction to obtain the average listening level without distortion yields:
6 dB 6 dB 6 dB 6 dB 6 dB -

15 dB = -9 dB (15 dB crest factor)
12 dB = -6 dB (12 dB crest factor)
9 dB = -3 dB (9 dB crest factor)
6 dB = 0 dB (6 dB crest factor)
3 dB = 3 dB (3 dB crest factor)

Converting dB back into watts:
Pw = 1oPdB/10xPref
= 63 mW (18 dB crest factor)

= 125 mW (15 dB crest factor)
= 250 mW (9 dB crest factor)
= 500 mW (6 dB crest factor)
= 1000 mW (3 dB crest factor)
= 2000 mW (15 dB crest factor)

~TEXAS

INSTRUMENTS
POST OFFICE SOX 655303 • DAlLAS. TEXAS 75265

(10)

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287-NOVEMBER 1999

APPLICATION INFORMATION
This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst case, which is 2 W of continuous power output with a 3 dB crest
factor, against 12 dB and 15 dB applications drastically affects maximum ambient temperature ratings for the
system. USing the power dissipation curves for a 5-V, 3-'1 system, the internal dissipation in the TPA0242 and
maximum ambient temperatures is shown in Table 4.
Table 4. TPA0242 Power Rating, S-V, 3-0., Stereo
PEAK OUTPUT POWER
(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE
-3°C

4

2W(3dB)

1.7

4

1000 mW (6 dB)

1.6

6°C

4

500 mW (9 dB)

1.4

24°C

4

250 mW (12 dB)

1.1

51°C

4

125 mW (15 dB)

0.8

78°C

4

63mW (18 dB)

0.6

96°C

Table 5. TPA0242 Power Rating, SOV, &-0., Stereo
POWER DISSIPATION
(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE

1250 mW (3 dB crest factor)

0.55

100°C

1000 mW (4 dB crest factor)

0.62

94°C

2.5W

500 mW (7 dB crest factor)

0.59

97°C

2.5W

250 mW (10 dB crest factor)

0.53

10~e

PEAK OUTPUT POWER

AVERAGE OUTPUT POWER

2.5W
2.5W

The maximum dissipated power, POmax, is reached at a much lower output power level for an 8 '1 load than for
a 3 '1 load. As a result, this simple formula for calculating POmax may be used for an 8 '1 application:
2V50
POmax = n;2R

(11 )

L

However, in the case of a 3 '1 load, the POmax occurs at a point well above the normal operating power level.
The amplifier may therefore be operated at a higher ambient temperature than required by the POmax formula
for a 3 '1 load.
The maximum ambient temperature depends on the heat sinking ability of the PCB system. The derating factor
for the PWP package is shown in the dissipation rating table (see page 4). Converting this to 8JA:

e

JA

=

1
= _1_
Derating Factor
0.022

= 450C/W

(12)

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

3-699

TPA0242
STEREO 2-W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SL0S287 - NOVEMBER 1999

APPLICATION INFORMATION
To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given 'E>JA' the maximum
allowable junction temperature, and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA0242 is
150·C. The internal dissipation figures are taken from the Power Dissipation vs Output Power graphs.
TA Max = T J Max - ElJA Po

(13)

= 150 - 45(0.6 x 2) = 96°C (15 dB crest factor)
NOTE:

Internal dissipation of 0.6 W is estimated for a 2-W system with 15 dB crest factor per channel.
Tables 4 and 5 show that for some applications no airflow is required to keep junction temperatures in the
specified range. The TPA0242 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150·C to prevent damage to the IC. Tables 4 and 5 were calculated for maximum
listening volume without distortion. When the output level is reduced the numbers in the table change
significantly. Also, using 8-n speakers dramatically increases the thermal performance by increasing amplifier
efficiency.

SElBTL operation
The ability of the TPA0242 to easily switch between BTL and SE modes is one of its most important cost saving
features. This feature eliminates the requirement for an additional headphone amplifier in applications where
internal stereo speakers are driven in BTL mode but external headphone or speakers must be accommodated.
Internal to the TPA0242, two separate amplifiers drive OUT+ and OUT-. The SElBTL input (terminal 15)
controls the operation of the follower amplifier that drives LOUT-and ROUT- (terminals 9 and 16). When
SElBTL is held low, the amplifier is on and the TPA0242 is in the BTL mode. When SElBTL is held high, the OUTamplifiers are in a high output impedance state, which configures the TPA0242 as an SE driver from LOUT+
and ROUT+ (terminals 4 and 21). 100 is reduced by approximately one-half in SE mode. Control of the SElBTL
input can be from a logic-level CMOS source or, more typically, from a resistor divider network as shown in
Figure 36.
20

RHPlN

23

RLiNEIN

R

8

MUX
ROUT+

21

....-+--Ul

RIN

Voo

ROUT-

COUTR

330IlF

16

100 Idl
SElBTL

15 100 Idl

~

n

r---~
Figure 36. TPA0242 Resistor Divider Network Circuit

~1ExAs

3-700

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

TPA0242
STEREO 2·W AUDIO POWER AMPLIFIER
WITH DC VOLUME CONTROL AND MUX CONTROL
SLOS287 - NOVEMBER 1999

APPLICATION INFORMATION
Using a readily available 1/8-in. (3.5 mm) stereo headphone jack, the control switch is closed when no plug is
inserted. When closed the 1OO-kQ/1-kQ divider pulls the SElBTL input low. When a plug is inserted, the 1-kQ
resistor is disconnected and the SE/BTL input is pulled high. When the input goes high, the OUT-amplifier is
shut down causing the speaker to mute (virtually open-circuits the speaker). The OUT+ amplifier then drives
through the output capacitor (Co) into the headphone jack.

PC BEEP operation
The PC BEEP input allows a system beep to be sent directly from a computer through the amplifier to the
speakers with few external components. The input is activated automatically. When the PC BEEP input is active,
both of the L1NEIN and HPIN inputs are deselected and both the left and right channels are driven in BTL mode
with the signal from PC BEEP. The gain from the PC BEEP input to the speakers is fixed at 0.3 VN and is
independent of the volume setting. When the PC BEEP input is deselected, the amplifier will return to the
previous operating mode and volume setting. Furthermore, if the amplifier is in shutdown mode, activating PC
BEEP will take the device out of shutdown and output the PC BEEP Signal, then return the amplifier to shutdown
mode.
When PCB ENABLE is held low, the amplifier will automatically switch to PC BEEP mode after detecting a valid
signal at the PC BEEP input. The preferred input signal is a square wave or pulse train with an amplitude of 1
V or greater. To be accurately detected, the signal must have a minimum of 1 VPP amplitude, rise and fall times
o less than 0.1 JlS and a minimum of 8 rising edges. When the signal is no longer detected, the amplifier will
return to its previous operating mode and volume setting.

fp

If it is desired to ac-couple the PC BEEP input, the value of the coupling capacitor should be chosen to satisfy
the following equation:

C

>

PCB - 2n

f pCB1(100

(14)

kQ)

The PC BEEP input can also be dc-coupled to avoid using this coupling capacitor. The pin normally sits at midrail
when no Signal is present.

shutdown modes
The TPA0242 employs a shutdown mode of operation designed to reduce supply current, 100, to the absolute
minimum level during periods of ,Ion use for battery-power conservation. The SHUTDOWN input terminal
should be held high during normal operation when the amplifier is in use. Pulling SHUTDOWN low causes the
outputs to mute and the amplifier to enter a low-current state, 100 = 150 ~A. SHUTDOWN should never be left
unconnected because amplifier operation would be unpredictable.

Table 6. HPILINE, SElBTL, and Shutdown Functions
AMPLIFIER STATE

INPUTSt
HPILINE

SElBTL

SHUTDOWN

INPUT

OUTPUT

X

X

Low

X

Mute

Low

Low

High

Line

BTL

Low

High

High

Line

SE

High

Low

High

HP

BTL

High

High

High

HP

SE

t Inputs should never be left unconnected.
X do not care

=

-!!1

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-701

3-702

TPA0243
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
DGQPACKAGE

• Ideal for Notebook Computers, PDAs, and
Other Small Portable Audio Devices
• 2 W Into 4-0 From 5-V Supply
• 0.6 W Into 4-0 From 3-V Supply

(TOP VIEW)
FILT_CAP
SHUTDOWN

• Stereo Head Phone Drive
• Mono (BTL) Signal Created by Summing
Left and Right Signals

VDD

BYPASS

RIN

LOIMO

LIN
GND
SRIMN
ROIMO

• Wide Power Supply Compatibility
3Vt05V
• Meets PC99 Desktop Specs (target)
• Low Supply Current
- 10 mA Typical at 5 V
- 9 mA Typical at 3 V
• Shutdown Control ••• 1 J1A Typical
• Shutdown Pin Is TTL Compatible
• -40°C to 85°C Operating Temperature
Range
• Space-Saving, Thermally-Enhanced MSOP
Packaging

description
The TPA0243 is a 2-W mono bridge-tied-Ioad (BTL) amplifier designed to drive speakers with as low as 4-0
impedance. The mono signal is created by summing left and right inputs. The amplifier can be reconfigured
on-the-fly to drive two stereo single-ended (SE) signals into head phones. This makes the device ideal for use
in small notebook computers, PDAs, digital personal audio players, anyplace a mono speaker and stereo head
phones are required. From a 5-V supply, the TPA0243 can delivery 2-W of power into a 4-0 speaker.
The gain of the input stage is set by the user-selected input resistor and a 50-kQ internal feedback resistor
(Av=- RFt RI). The power stage is intemallyconfigured with again of-1.25 VNin SE mode, and-2.5 VN in
BTL mode. Thus, the overall gain of the amplifier is 62.5 kW RI in SE mode and 125 kW RI in BTL mode. The
input terminals are high-impedance CMOS inputs, and can be used as summing nodes.
The TPA0243 is available in the 10-pin thermally-enhanced MSOP package (DGQ) and operates over an
ambient temperature range of -40°C to 85°C.

A

~

Please be aware that an important notice concemlng availability. standard warranty. and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incowrated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAU.AS. TEXAS 75265

Copyright @ 2000, Texas Instruments Incorporated

3-703

TPA0243
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE

SLOS279A - JANUARY 2000 - REVISED MARCH 2000

4

r------------- BvPAss--------1
VDD

3

1

VDD

ri
I II

FlLTCAP

I
I
I
I
I
I

BYPASS

50kn

1.25*R

1100kn

I

6

STIMN

I
I
I
I
I
I
I
I
I
I
I
I
I

7

CI
II---'\,R",I\r-'

50kn
StereoIMono

Control
50kn

Inpm

1.25*R

CI

9 LIN
Ir-~RNI~--~~~;

From
System Control

BYPASS
SHUTDOWN

-=

Cc

LOJMG- 1 10

I
I
I
I
I

21

Cc

ROJMO+

BYPASS

Left
Audio

VDD

I
I

I
51 RIN

Right
Audio
Input

8

GND

I
1I

Shmdown
andDepop
Circuitry

L _________________________

I
I
I
I
I
I
I
I
I
I

~

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

MSOpt
(DGQ)

-40·C to 85·C

TPA0243DGQ

MSOP
SYMBOLIZATION
AEK

t The DGQ package are available taped and reeled. To order a taped and reeled part, add the
suffix R to the part number (e.g., TPA0243DGQR).

~lExAs

3-704

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

1 kn

TPA0243
2·W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S279A - JANUARY 2000 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NO.
NAME
MONO-IN

110

DESCRIPTION

1

I

Mono input terminal

SHUTOOWN

2

I

SHUTOOWN places the entire device in shutdown mode when held low. TTL compatible input.

VOO
BYPASS

3

I

VOO is the supply voltage terminal.

4

I

BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected
to a O. 1-J,lF to 1-J.1F capacitor.
Right-channel input terminal

RIN

5

I

ROIMO

6

0

Right-output in SE mode and mono positive output in BTL mode

SRIMN

7

I

Selects between Stereo and Mono mode. When held high, the amplifier is in SE stereo mode, while held
low, the amplifier is in BTL mono mode.

GNO
LIN

8
9

I

Left-channel input terminal

LOIMO

10

0

Left-output in SE mode and mono negative output in BTL mode.

Ground terminal

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to VOO +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds ............................... 260°C
§ Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indlceted under "recommended operating conditions' is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
DISSIPATION RATING TABLE
PACKAGE
OGQ

DERATING FACTOR
2.14 w'II

17.1 mW/"C

1.37W

1.11 W

11 Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report
(literature number SLMAOO2), for more information on the PowerPAO peckage. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO
High-level input voltage, VIH

STIMN

ST/MN

MAX

2.5

5.5

IVoo:3V

2.7

I VOO:5V

4.5

SHUTDOWN
Low-level input voltage, VIL

MIN

UNIT
V
V

2
I VOO:3V

1.65

I VOO:5V

2.75

Operating free-air temperature, TA

V

0.8

SHUTOOWN

-40

85

°C

~TEXAS

INSTRUMENTS

POST OFFICE eox 655303 • OALlAS, TEXAS 75265

3-705

TPA0243
2-W MONO AUDIO POWER AMPLIFIER
WITH HEADPHONE DRIVE
SL0S279A - JANUARY 2000 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, Voo = 3 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDITIONS

IVOOI

Output offset veltage (measured differentially)

100

Supply current

IDD(SD)

Supply current, shutdewn mode

operating characteristics, Voo

MIN

TVP

MAX

UNIT

30

mV

9

14

rnA

1

10

pA

TVP

MAX

=3 V, TA =25°C, RL =4 n

PARAMETER

TEST CONDmONS
THO = 1%,

BTL mode

THD=O.I%,

SEmode,

Po

Output pewer, see Nete 1

THD+N

Tetal harmenic distortien plus neise

Po = 500 mW,

1=20 Hz to' 20 kHz

BOM

Maximum eutput power bandwidth

Gain=2,

THD=2%

MIN

660

mW

34

RL=320

UNIT

0.3%
kHz

20

NOTE 1: Output pewer is measured at the eutput terminals 0'1 the device at 1 = 1 kHz.

electrical characteristics at specified free-air temperature, Voo
noted)
PARAMETER

TEST CONDITIONS

IVOOI

Output effset veltage (measured. differentially)

100

Supply current

IDD(SD)

Supply current, shutdewn mede

operating characteristics, Voo

MIN

TVP

MAX

UNIT

30

mV

10

14

rnA

1

10

pA

=5 V, TA =25°C, RL =4 n
TEST CONDITIONS

PARAMETER
THO = 1%,

BTLmede

THD=O.I%,

SEmede,

Po

Output power, see Nete 1

THD+N

Tetal harmonic distertien plus
neise

PO=1 W,

1=20 Hz to' 20 kHz

BOM

Maximum eutput power bandwidth

Gain =2.5,

THO =2%

MIN

RL=320

NOTE 1: Output pewer is measured at the eutput terminals 0'1 the device at 1 = 1 kHz.

~TEXAS

INSTRUMENTS
3-706

=5 V, TA =25°C (unless otherwise

POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

TVP

MAX

UNIT

2

W

95

mW

0.2%
20

kHz

TPA1517
S-W STEREO AUDIO POWER AMPLIFIER
• TDA1517P Compatible
• High Power Outputs (6 W/Channel)
• Surface Mount Availability
20-Pin Thermal SOIC PowerPADTM

•
•
•
•

Thermal Protection
Fixed Gain ... 20 dB
Mute and Standby Operation
Supply Range ... 9.5 V -18 V
DWPPACKAGE
(TOP VIEW)

NEPACKAGE
(TOP VIEW)
IN1

GNDIHS

SGND

GNDIHS

SVRR

GND/HS

OUT1

GNDIHS

PGND

GNDIHS
GNDIHS

Vee
M/SB

GND/HS
GNDIHS

IN2

GND/HS

GND/HS

GND/HS

20
19
18
17
16
15
14
13
12
11

10
2
3
4
5
6
7
8
9
10

GND/HS
IN1
NC
SGND
SVRR
NC
OUT1
OUT1
PGND
GND/HS

(

GND/HS
IN2
NC
M/SB
Vec
NC
OUT2
OUT2
PGND
GND/HS

hL

11

Cross Section View Showing PowerPAD
NC - No internal connection

description
The TPA 1517 is a stereo audio power amplifier that contains two identical amplifiers capable of delivering 6 W
per channel of continuous average power into a 4-0 load at 10% THD+N or 5 W per channel at 1% THD+N.
The gain of each channel is fixed at 20 dB. The amplifier features a mute/standby function for power-sensitive
applications. The amplifier is available in Texas Instruments patented PowerPAD 20-pin surface-mount
thermally-enhanced package (DWP) that reduces board space and facilitates automated assembly while
maintaining exceptional thermal characteristics. It is also available in the 20-pin thermally enhanced DIP
package (NE).
AVAILABLE OPTIONS
PACKAGED DEVICES
TA

-40°C to 85°C

THERMALLY ENHANCED
PLASTIC DIP

THERMALLyt ENHANCED
SURFACE MOUNT
(DWP)

TPA1517NE

TPA1517DWP

tThe DWP package IS available taped and reeled. To order a taped and reeled part,
add the suffix R (e.g., TPAI517DWPR) .

.A.

~

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademark of Texas Instruments Incorporated.

-!!1

Copyright © 2000, Texas Instruments Incorporated

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

3-707

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

Terminal Functions
NAME

TERMINAL
DWP
NO.

NE
NO.

I/O

DESCRIPTION

IN1

2

1

I

IN1 is the audio input for channel 1.

SGND

4

2

I

SGND is the Input signal ground reference.

SVRR

5

3

OUT1

7,8

4

PGND

9, 12

5

OUT2

13,14

VCC
M/SB

IN2
GND/HS

SVRR Is the midrail bypass mode enable.

0

OUT1 Is the audio output for channel 1.

6

0

OUT2 is the audio output for channel 2.

16

7

I

VCC is the supply voltage input.

17

8

I

MISS Is the mute/standby mode enable. When held at less than 2 V, this signal enables the TPA1517
for standby operation. When held between 3.4 V and 8.8 V, this signal enables the TPA1517 for mute
operation. When held above 9.2 V, the TPA1517 operates normally.

19

9

I

1,10,
11,20

10-20

PGNO Is the power ground reference.

IN2 in the audio input for channel 2.
GNOIHS are the ground and heatslnk connections. All GNOIHS terminals are connected directly to
the mount pad for thermal-enhanced operation.

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)t
Supply voltage, Vee ...................................................................... 22 V
Input voltage, VI (IN1,.IN2) ................................................................. 22 V
Continuous total power dissipation .................... Internally limited (See Dissipation Rating Table)
Operating free-air temperature range, TA ........................................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg .................................................. -65°C to 150°C
Lead temperature 1,6 mm (1/16 inch) from case for 10 seconds: DWP or NE package ..........•. 260°C
t Stresses beyond those listed under "absolute maximum ratings' may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under "recommended operating conditions· is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: These devices have been classified as Class 1 ESO sensitive products per MIL-PRF-38535 Method 3015.7. Appropriate precautions
should be taken to prevent serious damage to the device.
DISSIPATION RATING TABLE
PACKAGE

TAS25°C

DERATING FACTOR

TA=70°C

TA=85OC

DWP:j:

2.94W

23.5mWfOC

1.88W

1.53W

NE:j:

2.85W

22.8mWfOC

1.82W

1.48W

:j: See the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Repotf
(literature number SLMAOO2), for more Information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments
Recommended Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
MIN
Supply VOltage, Vce
Operating free-air temperature, TA

~1ExAs

3-708

INSTRUMENTS
POST OFFICE BOX 855300 • DAllAS, TEXAS 75285

NOM

MAX

UNIT

9.5

18

V

-40

85

°e

TPA1517
S-W STEREO AUDIO POWER AMPLIFIER
SLOSI62B - MARCH 1997 - REVISED MARCH 2000

electrical characteristics,

Vee =12 V, TA =25°C (unless otherwise noted)

PARAMETER
ICC

Supply current

VO(DC)

DC output voltage

VIM/SB)

MlSB on voltage

VO(M)

Mute output voltage

ICC(SB)

Supply current in standby mode

TEST CONDITIONS

MIN

TYP

MAX

45

70

4

See Note 2

mA
V

9.5

V

2

VI=l V (max)

UNIT

mV

7

100

TYP

MAX

50

80

I!A

NOTE 2: At 6 V < VCC < 18 V the DC output voltage is approximately Vcd2.

electrical characteristics,

Vee = 14.5 V, TA = 25°C (unless otherwise noted)

PARAMETER

TEST CONDITIONS

ICC

Supply current

VOIDC)

DC output voltage

V(MlSB)

Voltage on MlSB terminal for normal operation

VO(M)

Mute output voltage

ICCISB)

Supply current in standby mode

MIN

See Note 2

9.5

V
mV

2
7

mA
V

5

VI= 1 V (max)

UNIT

100

I!A

NOTE 2: At 6 V < VCC < 18 V the DC output voltage is approximately Vcd2.

operating characteristic,

Vee = 12 V, RL = 4 n, f = 1 kHz, TA = 25°C

PARAMETER
Po

Output power (see Note 3)

SNR

Signal-ta-noise ratio

THO

Total harmonic distortion

IO(SM)

Non-repetitive peak output current

IOIRM)

Repetitive peak output current

TEST CONDITIONS

THD=10%

6

-3 dB
-1 dB

Supply ripple rejection ratio

M1SB=On,

Vn

Noise output voltage (see Note 4)
Channel separation

MAX

RL=8n,

f=lkHz

UNIT
W

84
PO=lW,

Low-frequency roll-off

Input impedance

TYP
3

High-frequency roll-off

ZI

. MIN

THO = 0.2%

dB

0.1%
4

A

2.5

A

45

Hz
kHz

20
f = 1 kHz

65

dB

60

kn

Rs=O,

MlSB=On

50

I1V(rms)

Rs =10kn,

M1SB=On

70

I1V(rms)

M/SB=Mute

50

I1V(rms)

Rs=10kn

58

Gain

18.5

Channel balance

dB

20

21

0.1

1

dB

NOTES: 3. Output power is measured at the output terminals of the IC.
4. Noise voltage is measured in a bandwidth of 20 Hz to 20 kHz.

~TEXAS

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3-709

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

operating characteristic,

Vee =14.5 V, RL =40., f = 1 kHz, TA =25°C
TEST CONDITIONS

PARAMETER
Po

Output power (see Note 3)

SNR

Signal-to-noise ratio

THO

Total harmonic dislortion

IO(SM)

Non-repetitive peak output current

IO~RMl

Repetitive peak output current

W

84

dB

Supply ripple rejection ratio

MlSB=On

Channel separation

W

0.1%

PO=1 W

-1 dB

Noise output voltage (see Note 4)

4

A

2.5

A

45

Hz

20

kHz

65

dB

60

k.Q

Rs=O,

MlSB=On

50

ILV(rms)

Rs = 10 k.Q,

MlSB=On

70

ILV(rms)

MlSB=Mute

50

ILV(rms)

Rs=10k.Q

58
18.5

Gain
Channel balance

dB

20

21

dB

0.1

1

dB

NOTES: 3. Output power is measured at the output terminals of the IC.
4. Noise voltage is measured in a bandwidth of 22 Hz to 22 kHz.

TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
ICC

THO+N

Supply current

vs Supply voltage

Power supply rejection ratio

vs Frequency

1
2,3

VCC=12V

vs Frequency
vs Power output

4,5,6
10,11

VCC = 14.5 V

vs Frequency
vs Power output

7,8,9
12,13

Total harmonic distortion plus noise

Crosstalk

vs Frequency

14,15

Gain

vs Frequency

16

Phase

vs Frequency

16

Vn

Noise voltage

vs Frequency

17,18

Po

Output power

vs Supply voltage
vs Load resistance

Po

Power dissipation

vs Output power

~lExAs

INSTRUMENTS
3-710

UNIT

6

-3 dB

Vn

MAX

THO < 10%

High-frequency roll-off
Input Impedance

TYP
4.5

Low-frequency roll-off

ZI

MIN

THO =0.2%

POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

19
20
21,22

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS1628 - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
SUPPLY RIPPLE REJECTION RATIO

SUPPLY CURRENT

vs

vs

SUPPLY VOLTAGE

FREQUENCY
0

100

I

III

75

""E
'E
~:::I
:::I

I/)

I

0

E

- --

--- -- -

50

~
a.

-20

~c
~

-30

-40

'iii

-50

GI

-60

0

I

0

"I

GI

II:

Q.

a.

I

I

VCC=12V
RL=4Q
CB=looj.1F

-10

I

t--

I--

Ii: -70

~
a.

25

:::I

I/)

-80

-90

o

10

8

12
14
16
VCC - Supply Voltage - V

18

-100
100

20

Figure 1

Figure 2
TOTAL HARMONIC DISTORTION PLUS NOISE

SUPPLY RIPPLE REJECTION RATIO

o
-10

vs

vs

FREQUENCY

FREQUENCY
10%

V~=1~.5~ I
-

.!!!

RL=4Q

0

z

-20

+

c

i

-30

E

S

J

-40

~

.!!

-60

8:

VCC=12V
RL=4Q
PO=3W
Both Channels

GI

III

"I

10k

1k
f - Frequency - Hz

0
0

1%

..

-50

..........

Ii: -70

J

0

-

fi

--

"...,.

:J:

!i

0.1%

""

~
I

-80

Z

-90

I-

-100
100

7'

';:

+
Q

:J:

0.01%
1k
f - Frequency - Hz

10K

20

Figure 3

100
1k
f - Frequency - Hz

10 k 20k

Figure 4

~TEXAS

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3-711

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY

FREQUENCY

vs

10%

10%
VCC=12V
RL=SQ
PO=1W
Both Channels

·1z
+

c

z

+

c

0

i:

~.5!

0

i:

1%

~

c

I

0

!
:l!

li

VCe=12V
RL=32Q
PO=0.25W

Jo·

-

~I-"

0.1%

~
I

1%

~

I
j

~
~

0.1%

I

z+

Z

+
Q

Q

:c

:c

I-

I-

0.01%

0.01%
20

100

1k
f - Frequency - Hz

10k 20k

20

Figure 6

TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs
FREQUENCY

FREQUENCY

vs

10%

10%
VeC=14.5V
RL=4Q
PO=3W

••
Z
+

~

Vee = 14.5 V
RL=SQ
PO=1.5W

J0
z

+

c
0

i:

1%

~

is
u

c0

Ili

10k 20k

f - Frequency - Hz

FigureS

~

1k

100

,

~

I

~

V

0.1%

1%

~

j

Z
+

z+

:c

j!:

I

-

,~

0.1%

I

Q

Q

I-

0.01%

0.01%
20

100

1k
f - Frequency - Hz

10k 20k

20

1k
f - Frequency - Hz

Figure 7

3-712

100

FigureS

~TEXAS .
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10k 20k

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

FREQUENCY

POWER OUTPUT

10%

.!!
0

z

10%

..
z

Vee = 14.5 V
RL=320
Po=0.25W

CD

Vee = 12 V
RL=40
Both Channels

CD

15

+
c

-

+

c

0

0

'E0

~

1%

~

1%

~

..

..

c

!

~

:!

';'

S

c0

Ii

f=20Hz

J:

S

0.1%

{!.

0.1%

{!.

I

Z

I
Z

Q

Q

+

t- r--

I-

20

100

1k

0.01%
0.01

10 k 20 k

0.1
Po - Power Output - W

f - Frequency - Hz

TOTAL HARMONIC DISTORTION PLUS NOISE

10%

=

15
z
+
c

TOTAL HARMONIC DISTORTION PLUS NOISE

vs

vs

POWER OUTPUT

POWER OUTPUT
10%

t:::: Vee=12V
f= RL=SO
~ Both Channels

.~

~ Both Channels

z

r-- t-

0

'E0

f=20kHz

is

1ii
is

c0

'2

..

{!.

F Vec=14.5V

I- RL=40

+
c

1%

1%

.
0

f=2OHz
0.1%

-'I

i

r-

J:

./

Oi

0.1%

;2

f= 1 kHz

I

10

Figure 10

Figure 9

IS

~

J:

0.01%

;

tHm..
f = 1 kHz

+

j!:

~

f= 20 kHz

t-

-..

f=20kHz

f=20Hz

IT ----~

f=1kHz

I

Z

Z

+
Q

+
Q

J:

J:

I-

I-

0.01%
0.01

0.1
Po - Power Output - W

10

0.01%
0.01

Figure 11

0.1
Po - Power Output - W

10

Figure 12

~TEXAS

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3-713

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
TOTAL HARMONIC DISTORTION PLUS NOISE

10%

CROSSTALK

va

va

POWER OUTPUT

FREQUENCY
-40

F

VCC=14.5V
RL=SCl
r- Both Channels

i=

I

~

VCC=1'2'V
RL=4Cl
PO=3W
Both Channels

-45

r-

+
c

-50

0

:e

f= 20 kHz

1%

i

ID

'1:1
I

~0

~

I

1111

fl=I4clI~z
-r-t-I4.

!§
01
:r

i

0.1%

~

.......

~

(.I

~~

:::;:..-

~

-60
-65
-70

f=l kHz

I
Z

-55

+

Q

-75

.-:r
0.01%
0.01

-80
0.1
Po - Power Output -

10

100

20

w

lk
f - Frequency - Hz

Figure 13

Figure 14
CROSSTALK

va
FREQUENCY
-40
VCC=14.5V
RL=4Cl
PO=5W
Both Channels

-45
-50

~
ID

'1:1
I

1
S

-55

...
:;.....-

~

-60
-65
-70
-75

-80

20

100

10k 20k

1k
f - Frequency - Hz

Figure 15

~TEXAS

3-714

INSTRUMENTS
POST OFFICE BOX 655303 •

DAllAS~

TEXAS 75265

10k 20k

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
GAIN AND PHASE

vs
FREQUENCY
20

II~

~
10

~

-

1-

\

0
ID

"cI

'ii

r-..

-10

Phase

ClI

2000

VCC=12V
RL=40

1000

I'

,

-20

-

-

1-

'\

-30

-1000

~

-40
10

100

1k
10k
f - Frequency - Hz

100k

-2000

1M

Figure 16
NOISE VOLTAGE

NOISE VOLTAGE

vs

vs

FREQUENCY

FREQUENCY
Vcc = 14.5 V
BW = 22 Hz to 22 kHz
RL=40
Both Channels

VCC= 12 V
BW = 22 Hz to 22 kHz
RL=40
Both Channels

~

~

I

t
~

:

I

t
I

0.1

0.1

~

"0

z
I

c

I

>

C

>

0.01

20

100

1k
f - Frequency - Hz

10 k 20 k

0.01

20

100

1k

10k 20 k

f - Frequency - Hz

Figure 17

Figure 18

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3-715

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

TYPICAL CHARACTERISTICS
OUTPUT POWER

OUTPUT POWER

vs

vs

SUPPLY VOLTAGE

LOAD RESISTANCE

8

6
THDcl%

V

6

R~

==I

I

V

0

I

,p

./

2

o

V
./

4

'5
Co
'5

THDcl%

V

~V
8

9

,/

.........

10

,/

RL=V ~

....V

5

~

III

4

I

\

\Jee =1 4.5V

==I

I,p

3

I

2

17

I

'
\ \
K .....
"-

o

18

1

I

II
\ f\ IVee=12V
\

/1"""

11 12 13 14 15 16
Vee - Supply Voltage - V

l

I

""
.......

POWER DISSIPATION

vs

OUTPUT POWER

OUTPUT POWER
3.5

I

3

==cI

2.5

.B-

is

I

2

1.5

/

i

......... ~=4n

~

(

is

2

Do

1.5

!
I

V::-l"::'
0.5

o

2.5

"ii

I

~

/

I
V

............

RL=4n

~

RL~

~

0.5
2
3
4
Po - Output Power - W

5

6

o

Figure 21

4
2
3
Po - Outpul Power - W

Figure 22

~TEXAS

3-716

---

Vee = 1 /

3

i

---

vs

Vee=12V

0

-

t- r-

r- r-

Figure 20

POWER DISSIPATION

==cI

.......

2 4 6 8 10 12 14 16 18 20 22 24 26 28 3032
RL - Load Resistance - n

Figure 19

3.5

.......

INSTRUMENTS
POST OFFICE BOX 65S303 • DALlAS, TEXAS 75265

5

6

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

APPLICATION INFORMATION

amplifier operation
The TPA1517 is a stereo audio power amplifier designed to drive 4-0 speakers at up to 6 W per channel.
Figure 23 is a schematic diagram of the minimum recommended configuration of the amplifier. Gain is internally
fixed at 20 dB (gain of 10 VN).
vee

7

elR

Right

Vee
eST 1 jJ.F

---1 f-----'.1+-"IN-'.21~_ _-I
1 jJ.F
OUT1

4

2.1 Vref
Vee
2 SGND

Ref

Vee
15kQ

-=-

Mute
Standby

3 SVRR

MlSB 8

15kQ

eBT
2.21!F -=-

Sl

-=-

Mute/Standby Switch
(see Note A)

18kQ
MutelStandby Select
(see Note B)

ell
Left

----j f----,9,¥,IN!!o.2+---1
11!F

GND/HS
10-20
eopper Plane

NOTES: A. When 51 is open, the TPAI517 operates normally. When this switch is closed, the device is in mute/standby mode.
B. When 52 is open, activating 51 places the TPA1517 in mute mode. When 52 is closed, activating SI places the TPA1517 in standby
mode.
C. The terminal numbers are for the 20-pin NE package.

Figure 23. TPA1517 Minimum Configuration
The following equation is used to relate gain in VN to dB:
G dB

=

IV)

20 LOG( G v

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TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
The audio outputs are biased to a mid rail voltage which is shown by the following equation:
V MID =

Vee
--r

The audio inputs are always biased to 2.1 V when in mute or normal mode. Any dc offset between the input signal
source and the input terminal is amplified and can seriously degrade the performance of the amplifier. For this
reason, it is recommended that the inputs always be connected through a series capacitor (ac coupled). The
power outputs, also having a dc bias, must be connected to the speakers via series capacitors.
mute/standby operation

The TPA1517 has three modes of operation; normal, mute, and standby. They are controlled by the voltage on
the MISS terminal as described in Figure 24. In normal mode, the TPA1517 amplifies the signal applied to the
two input terminals providing low impedance drive to speakers connected to the output terminals. In mute mode,
the amplifier retains all bias voltages and quiescent supply current levels but does not pass the input signal to
the output. In standby mode, the intemal bias generators and power-drive stages are turned off, thereby
reducing the supply current levels.
22

>
I

III

ig
j
~
Sa.
.5

9.2
8.8

I

ii

~

>

3.4
2
0

Figure 24. Standby, Mute, and Normal (On) Operating Conditions

The designer must take care to place the control voltages within the defined ranges for each desired mode,
whenever an external circuit is used to control the input voltage at the MISS terminal. The undefined area can
cause unpredictable performance and should be avoided. As the control voltage moves through the undefined
areas pop or click sounds may be heard in the speaker. Moving from mute to normal causes a very small click
sound. Whereas moving from standby to mute can cause a much larger pop sound. Figure 25 shows external
circuitry designed to help reduce transition pops when moving from standby mode to normal mode.

~TEXAS

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TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
Figure 25 is a reference schematic that provides TTL-level control of the M/SB terminal. A diode network is also
included which helps reduce turn-on pop noises. The diodes serve to drain the charge out of the output coupling
capacitors while the amplifier is in shutdown mode. When the M/SB voltage is in the normal operating range,
the diodes have no effect on the ac performance of the system.
VCC

7

CIR

Right

-1

VCC
CST 1 f.1F

1

IN1

-:::

1 f.1F

COR 47Of.1F
OUTl

1~S1JJ

4

-:::

18110
VCC
2 SGND

Ref

VCC
2kO

5 PGND

10kO
10110

15110

-:::

Mute
Standby

3 SVRR
15110

CBT
2.2f.1F _

M/SB 8

47110

47kO

2kO

TTL Control
low-Mute
High-On

lN914

-:::

10110

6.8kO

18kO
2.1 Vref

-:::

COL
OUT2 6
Cil
left

~
1 f.1F

4ro~
-:::

GND/HS
10-20
-:::

Copper Plane
NOTES: A. When Sl is closed, the depop circuitry is active during standby mode.
B. When S2 is open, activating SI places the TPA 1517 in mute mode. When S2 is closed, activating SI places the TPA 1517 in standby
mode.
C. The terminal numbers are for the 20-pin NE package.

Figure 25. TTL Control with POP Reduction

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3-719

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOSl62B - MARCH 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
component selection
Some of the general concems for selection of capacitors are:
•
•
•

Leakage currents on aluminum electrolytic capacitors
ESR (equivalent series resistance)
Temperature ratings

leakage currents
Leakage currents on most ceramic, polystyrene, and paper capacitors are negligible for this application.
Leakage currents for aluminum electrolytic and tantalum tend to be higher. This is especially important on the
input terminals and the SVRR capacitor. These nodes encounter from 3 V to 7 V, and need to have leakage
currents less than 1 IlA to keep from affecting the output power and noise performance.

equivalent series resistance
ESR is mainly important on the output coupling capacitor, where even 1 0 of ESR in Co with an 8-0 speaker
can reduce the output drive power by 12.5%. ESR should be considered across the frequency range of interest,
(I.e., 20 Hz to 20 kHz). The following equation calculates the amount of power lost in the coupling capacitor:
% Power in Co = E~R
L

In general, the power supply decoupling requires a very low ESR as weillo take advantage of the full output
drive current.

temperature range
The temperature range of the capacitors mayor may not seem like an obvious thing to specify, but it is very
import~nt. Many of the high-density capacitors perform very differently at different temperatures. When
consistent high performance is required from the system over temperature in terms of low THO, maximum
output power, and turn-on/off popping, then interactions of the coupling capacitors and the SVRR capacitors .
need to be considered, as well as the change in ESR on the output capacitor with temperature.

turn-on pop consideration
To select the proper input coupling capacitor, the designer should select a capacitor large enough to allow the
lowest desired frequency pass and small enough that the time constant is shorter than the output RC time
constant to minimize tum-on popping. The input time constant for the TPA1517 is determined by the input
60-kO resistance of the amplifier, and the input coupling capacitor according to the following generic equation:
T

1
C - 21tRC

For example, 8-0 speakers and 220-~F output coupling capacitors would yield a 90-Hz cut-off point for the
output RC network. The input network should be the same speed or faster ( > 90 Hz Tc). A good choice would
be 180 Hz. As the input resistance is 60 kO, a 14-nF input coupling capacitor would do.
The bypass-capacitor time constant should be much larger (><5) than either the input coupling capacitor time
constant or the output coupling capacitor time constants. In the previous example with the 220-~F output
coupling capacitor, the designer should want the bypass capacitor, TC, to be in the order of 18 Hz or lower. To
get an 18-Hz time constant, Cs is required to be 1 ~F or larger because the resistance this capacitor sees is
7.5kO.

="TEXAS
3-720

INSTRUMENTS

POST OFACE BOX es5303 -DALlAS. TEXAS 75265

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS162B - MARCH 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
In summary, follow one of the three simple relations presented below, depending on the tradeoffs between low
frequency response and turn-on pop. If depop performance is the top priority, then follow:
7500 C B > 5R LCo > 300000 C,
If low frequency ac response is more important but depop is still a consideration then follow:

1

2n:60000 C, < 10 Hz
Finally, if low frequency response is most important and depop is not a consideration then follow:

1

1

2n:60000 C,

S;

2n:RL C,

S;

flow

thermal applications
Linear power amplifiers dissipate a significant amount of heat in the package under normal operating conditions.
A typical music CD requires 12 dB to 15 dB of dynamic headroom to pass the loudest portions without distortion
as compared with the average power output. Figure 19 shows that when the TPA1517 is operating from a 12-V
supply into a 4-0 speaker that approximately 3.5 W peaks are possible. Converting watts to dB using the
following equation:
.

P dB

=

10Log

(:w)
ref

1

10L09(3 5)

=

5.44 dB

Subtracting dB for the headroom restriction to obtain the average listening level without distortion yields the
following:
5.44 dB - 15 dB = - 9.56 dB (15 dB headroom)
5.44 dB - 12 dB = - 6.56 dB (12 dB headroom)
Converting dB back into watts:
- 1OPdB/ 10
P
P wX
ref
= 111 mW (15 dB headroom)
= 221 mW (12 dB headroom)

This is valuable information to consider when attempting to estimate the heat dissipation requirements for the
amplifier system. Comparing the absolute worst cast, which Is 3.5 W of continuous power output with 0 dB of
headroom, against 12-dB and 15-dB applications drastically affects maximum ambient temperature ratings for
the system. Using the power dissipation curves for a 12-V, 4-0 system, internal dissipation in the TPA1517 and
maximum ambient temperatures are shown in Table 1.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

3-721

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOSl62B - MARCH 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
Table 1. TPA1517 Power Rating
PEAK OUTPUT POWER

(W)

AVERAGE OUTPUT POWER

POWER DISSIPATION

(W/Channel)

MAXIMUM AMBIENT
TEMPERATURE
-34°C

3.5

3.5W

2.1

3.5

1.nW(3dB)

2.4

-Sloe

3.5

884 mW (S dB)

2.25

-48°C

3.5

442mW(9dB)

1.75

-4°C

3.5

221 mW (12 dB)

1.5

18°C

3.5

111 mW (15 dB)

1.25

40°C

The maximum ambient temperature depends on the heatsinking ability of the PCB system. Using the derating
factor for the NE package with 4 square inches of copper area is 22.8 mW/oC and 38.8 mWrC respectively.
Converting this to 6JA:
Derating
For 0 CFM:

=_1_
0.0228

=

43.9°C;W

To calculate maximum ambient temperatures, first consider that the numbers from the dissipation graphs are
per channel so the dissipated heat needs to be doubled for two channel operation. Given 6JA, the maximum
allowable junction temperature and the total internal dissipation, the maximum ambient temperature can be
calculated with the following equation. The maximum recommended junction temperature for the TPA 1517 is
150°C.
T J Max - 6JA Po
150 - 43.9{1.25 x 2)

=

40°C (15 dB headroom, 0 CFM)

Table 1 clearly shows that for most applications some airflow is required to keep junction temperatures in the
specified range. The TPA1517 is designed with thermal protection that turns the device off when the junction
temperature surpasses 150°C to prevent damage to the IC. Using the DWP package on a multilayer PCB with
internal ground planes can achieve better thermal performance. Table 1 was calculated for a maximum volume
system; when the output level is reduced, the numbers in the table change significantly. Also using 8-0 speakers
dramatically increases the thermal performance by increasing amplifier efficiency.

~TEXAS

INSTRUMENTS

3--722

POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

TPA1517
6-W STEREO AUDIO POWER AMPLIFIER
SLOS1628 - MARCH 1997 - REVISED MARCH 2000

APPLICATION INFORMATION
TPA1517 NE THERMAL RESISTANCE, 9JA
vs
COPPER AREA

90
80 1\
70

~

0

\

\

60

"r-..

I

...

20
10

o

o

2

3

4

5

6

7

8

9

10

Copper Area - In2

Figure 26

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

3-723

3-724

4-1

Contents
Page
TPA0211

..

"tJ

oQ.
C

n
,..
"tJ

i

<

I

4-2

2-W Mono Audio Power Amplifier ............................. 4-3

TPA0211
2·W MONO AUDIO POWER AMPLIFIER
DGNPACKAGE

• Ideal for Wireless Communicators,
Notebook PCs, PDAs, and Other Small
Portable Audio Devices
• 2 W Into 4-0 From 5-V Supply

(TOP VIEW)

IN

Va-

""SH"'U"'T""D"'O"'W'""N",,...,r-I

• 0.6 W Into 4-0 From 3-V Supply

VDD

• Wide Power Supply Compatibility

BYPASS

GND
S8BTL

Vo+

3Vt05V
• Low Supply Current
- 8 mA Typical at 5 V
- 4 mA Typical at 3 V
• Shutdown Control ••• < 1 IlA Typical
• Shutdown Pin is TTL Compatible
• -4O°C to 85°C Operating Temperature
Range
• Space-Saving, Thermally-Enhanced MSOP
Packaging

~~

~

W

The TPA0211 is a 2-W mono bridge-tied-Ioad (BTL) amplifier designed to drive speakers with as low as 4-0
impedance. The device is ideal for use in small wireless communicators, notebook PCs, PDAs, anyplace a
mono speaker and stereo head phones are required. From a 5-V supply, the TPA0211 can delivery 2-W of power

:;:

~a~~~~

~

W

~

The gain of the input stage is set by the user-selected input resistor and a 50-kQ intemal feedback resistor
(Av =- RFt RI). The power stage is internally configured with a gain of -1.25 VN in,SE mode, and -2.5 VN in
BTL mode. Thus, the overall gain of the amplifier is 62.5 knI RI in SE mode and 125 knI RI in BTL mode. The
input terminals are high-impedance CMOS inputs, and can be used as summing nodes.

0

The TPA0211 is available in the B-pin thermally-enhanced MSOP package (DGN) and operates over an ambient
temperature range of -4O°C to B5°C.

0

I-

::)

C

~
~

.A

~

Please be aware that an important notice concemlng availability, standard warranty, and use In critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PowerPAD is a trademar\( of Texas Instruments Incorporated.

a::

PRODUCT PREVIEW Infonnatlan concerns products In the fannatlve Dr
Pheee of deveIopmenl Cheracterlatlc data and othar
•
DI18 are dee
JIOIII. 1Uaa Jnotrurnents '"""""" the rlg/ll fa
or

dJaoontlnue~... products wtthout notl...

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS. TEXAS 75265

Copyright © 2000, Texas Instruments Incorporated

TPA0211
2·W MONO AUDIO POWER AMPLIFIER
SLOS275A - JANUARY 2000 - REVISED MARCH 2000

4

1---------VDD

31
I
I
I
I
I
I
I
I

VDD

VDD

BYPASS

50kn

1.25*R

100kn

1 liN

3:
w

BYPASS

50kn

5>
w
a:
a.

StereoiMono
Control

50kn

SElBTL

1.25*R

b

:::)

VO-

c
o

a:
a.

BYPASS
From
System Control

2

SHUTDOWN

Shutdown
and Depop
CIrcuitry

L ______________________

I
I
I
I
I
I
I6
I
I
I
I
I
I
18

100kn
1 kn

I
I
I
I
I
I
I
I
I
I

~

AVAILABLE OPTIONS
PACKAGED DEVICES
TA

MSOpt
(DGN)

-40°C to 85°C

TPA0211DGN

MSOP
SYMBOLIZATION
AEG

tThe DGN package are available taped and reeled. To order a taped and reeled part, add the
suffix R to the part number (e.g., TPA0211DGNR).

~TEXAS

4-4

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

TPA0211
2-W MONO AUDIO POWER AMPLIFIER
SLOS275A - JANUARY 2000 - REVISED MARCH 2000

Terminal Functions
TERMINAL
NAME

NO.

1/0

DESCRIPTION

I

BYPASS is the tap to the voltage divider for internal mid-supply bias. This terminal should be connected to
a 0.1-I1F to 1-I1F capacitor.

BYPASS

4

GNO

7

IN

1

I

IN is the audio input terminal.

SElBTL

6

I

When SE/BTL is held low, the TPA0211 is in BTL mode. When SElBTL is held high, the TPA0211 is in SE
mode.

I

SHUTOOWN places the entire device in shutdown mode when held low. TTL compatible input.

GNO is the ground connection.

SHUTDOWN

2

VOO

3

VO+

5

0

VO+ is the positive output for BTL and SE modes.

Vo-

8

0

Vo- is the negative output in BTL mode and a high-impedance output in SE mode.

VOO is the supply voltage terminal.

absolute maximum ratings over operating free-air temperature range (unless otherwise noted)§
Supply voltage, Voo ....................................................................... 6 V
Input voltage, VI ............................................................ -0.3 V to Voo +0.3 V
Continuous total power dissipation ..................... internally limited (see Dissipation Rating Table)
Operating free-air temperature range, TA (see Table 3) ............................... -40°C to 85°C
Operating junction temperature range, TJ .......................................... -40°C to 150°C
Storage temperature range, Tstg •..••..••..••......••••....••...••.•...••••....•.. -65°C to 150°C
Lead temperature 1,6 mm (1116 inch) from case for 10 seconds ............................... 260°C

~

5>
w
a:

§ Stresses beyond those listed under "absolute maximum ratings" may cause permanent damage to the device. These are stress ratings only, and

D..

functional operation of the device at these or any other conditions' beyond those indicated under "recommended operating conditions" is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

I-

DISSIPATION RATING TABLE
PACKAGE
OGN

DERATING FACTOR
2.14 w'II

17.1 mW/oC

TA
1.37W

=85°C

1.11W

'\I Please see the Texas Instruments document, PowerPAD Thermally Enhanced Package Application Report

D..

(literature number SLMAOO2), for more information on the PowerPAO package. The thermal data was
measured on a PCB layout based on the information in the section entitled Texas Instruments Recommended
Board for PowerPAD on page 33 of the before mentioned document.

recommended operating conditions
Supply voltage, VOO

High-level input voltage, VIH

ST/MN

ST/MN

MAX

2.5

5.5

2.7

I VOO=5V

4.5

UNIT
V

V

2

SHUTOOWN

LOW-level input voltage, VIL

MIN

I VOO=3V

I VOO=3V

1.65

IVDD=5V

2.75

-40

Operating free-air temperature, TA

-!!1

TEXAS
INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS, TEXAS 75265

V

0.8

SHUTDOWN

o
::l
o
o
a:

85

°C

TPA0211
2·W MONO AUDIO POWER AMPLIFIER
SL0S275A - JANUARY 2000 - REVISED MARCH 2000

electrical characteristics at specified free-air temperature, VDD
noted)
PARAMETER

=3 V, TA =25°C (unless otherwise

TEST CONDmoNS

MIN

TYP

MAX

IVool

Output offset voltage (measured differentially)

100

Supply current

4

IOO(SO)

Supply current, shutdown mode

1

10

TYP

MAX

30

UNIT
mV
mA

IIA

operating characteristics, VDD = 3 V, TA = 25°C, RL = 4 Q
PARAMETER

TEST CONDmoNS
THO=1%,

BTL mode

THO=0.1%,

SEmode,

Po

Output power, see Note 1

THO+N

Total harmonic distortion plus nOise

Po = 500 mW,

f= 20 Hz to 20 kHz

BaM

Maximum output power bandwidth

Gain=2,

THO =2%

MIN

660

mW

33

RL=320

UNIT

0.3%
20

kHz

NOTE 1: Output power is measured at the output terminals of the device at f = 1 kHz.

;:
w

5>
w
a:
a.

t3

;:)

c
o
a:
a.

electrical characteristics at specified free-air temperature, VDD = 5 V, TA = 25°C (unless otherwise
noted)
PARAMETER

TEST CONDmONS

MIN

TYP

IVool

Output offset voltage (measured differentially)

100

Supply current

8

IOO(SO)

Supply current, shutdown mode

1

MAX
30

UNIT
mV
mA

10

IIA

operating characteristics, VDD = 5 V, TA = 25°C, RL = 4 Q
PARAMETER

TEST CONDITIONS
THO = 1%,

BTL mode

THO = 0.1%,

SEmode,

Po

Output power, see Note 1

THO+N

Total harmonic distortion plus
noise

PO= 1.5W,

1 = 20 Hz to 20 kHz

BOM

Maximum output power bandwidth

Gain = 2.5,

THO=2%

MIN
RL=320

NOTE 1: Output power is measured at the output terminals 01 the device at f = 1 kHz.

~TEXAS

INSTRUMENTS
POST OFFICE SOX 655303 • DALLAS, TEXAS 75265

TYP

MAX

UNIT

2

W

92

mW

0.2%
20

kHz

5-1

Contents
Page

»

Design Considerations for Class-D Audio Power Amplifiers
Application Report ................................................. 5-3
Mono Configuration of the TPA005D02 Class-D Audio Power Amplifier
Application Report ................................................ 5-31
PowerPAD Thermally Enhanced Package
Technical Brief .................................................... 5-39
Reducing and Eliminating the Class-D Output FiRer
Application Report ................................................ 5-85

"C
"C

-_.

n
m
,..

_.

o

:::J

:rJ

CD
"C

o

:3tn

5-2

Design Considerations for
Class-D Audio Power Amplifiers
Application Report

Literature Number: SLOA031
August 1999

~TEXAS

INSTRUMENTS

Printed on Recycled Paper

5-3

IMPORTANT NOTICE
Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue
any product or service without notice, and advise customers to obtain the latest version of relevant information
to verify, before placing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those
pertaining to warranty, patent infringement, and limitation of liability.

TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with TI's standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily
performed, except those mandated by government requirements.
CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
DEATH, PERSONAL INJURY, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE ("CRITICAL
APPLICATIONS"). TI SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, AUTHORIZED, OR
WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT DEVICES OR SYSTEMS OR OTHER
CRITICAL APPLICATIONS. INCLUSION OF TI PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO
BE FULLY AT THE CUSTOMER'S RISK.
In order to minimize risks associated with the customer's applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. TI's publication of information regarding any third
party's products or services does not constitute TI's approval, warranty or endorsement thereof.

Copyright © 1999, Texas Instruments Incorporated

5-4

Contents
1 Introduction . ................................................................................ 5-7
2 Class-D Amplifier Circuits .................................................................... 5-8
2.1 Input Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5-8
2.2 Output Circuit ......................................................................... 5-10
2.2.1 Filter Design ................................................................... 5-11
2.2.2 Design Example ............................................................... 5-13
2.2.3 Component Selection ........................................................... 5-14
2.3 Charge Pump Circuit .............................. , ...... '" ........................... 5-15
2.4 Switching Circuit ....................................................................... 5-16
3 Headphone Circuit .......................................................................... 5-18
4 Control and Indicator Circuits . ............................................................... 5-20
4.1 Shutdown ............................................................................. 5-20
4.2 Mute ................................................................................. 5-20
4.3 Mode ................................................................................ 5-20
4.4 Fault Indicators ........................................................................ 5-21
5 Device Power Supply Decoupling ............................................................ 5-22
5.1 Bulk Capacitors ....................................................................... 5-22
5.2 Small Decoupling Capacitors ............................................................ 5-24
6 PCB Layout . ................................................................................ 5-25
6.1 Ground Plane ......................................................................... 5-25
6.2 Power Plane .......................................................................... 5-26
6.3 Inputs and Outputs .................................................................... 5-27
6.4 General PowerPAD Considerations ...................................................... 5-27
7 References . ................................................................................ 5-29

Design Considerations for C/ass-D Audio Power Amplifiers

5-5

Figures

List of Figures
1 Class-D Input Circuit and Filter ................................................................... 5-8
2 Class-D Output Circuit and Filter ................................................................ 5-10
3 BTL Half-Circuit Model ......................................................................... 5-11
4 Combination of Two Half-Circuit Models .......................................................... 5-12
5 Complete BTL Output Filter ..................................................................... 5-14
6 Tripier Charge Pump Circuit ....................................................... " ............ 5-15
7 Switching Circuit for the Right Channel ........................................................... 5-16
8 Headphone Amplifier Circuit, Right Channel ........... " .. , '" ., .................................. 5-18
9 Mode Control Circuit Featuring Headphone Jack Control ........................................... 5-21
10 Class-D Power Bus, 48-Pin TSSOP Package .................................................... 5-22
11 Power Supply Bulk Decoupling Capacitor Circuit .......... , ....................................... 5-23
12 PowerPAD PCB Etch and Via Pattern ................................................ .' .......... 5-28

List of Tables
1 Second-Order Butterworth LCL Values .......................................................... 5-13
2 Audio Power Amplifier Subcircuit Ground Pins .................................................... 5-25

5-6

SLOA031

Design Considerations for C/ass-D Audio Power Amplifiers
Richard Palmer
ABSTRACT
This application report provides background information, general equations, and
component selection criteria for proper design and implementation of the Texas
Instruments class-D audio power amplifiers. Topics include class-D switching and charge
pump circuits, signal conditioning of the audio inputs and outputs for both the class-D and
class-AB headphone amplifiers, IC control and indicator circuits, power supply
decoupling, and PCB layout.

1 Introduction
Circuit design and layout plays a large role in creating or reducing distortion in
class-O audio power amplifiers. The high-frequency-switching characteristics
of class-O output stages offer some interesting design challenges over
conventional class-AS amplifiers. This application report provides the
background information necessary to properly design Texas Instruments (Tim)
class-O stereo audio power amplifiers into an audio solution.
Texas Instruments offers several class-O stereo audio power amplifiers, each of
which is featured on an evaluation module (EVM), available from TI. All
information appearing in this report originated from the design of the SLOP204
EVM, which features the TPA005014 class-O stereo audio power amplifier IC.
The TPA005014 EVM is capable of driving 2 W into a 4-(1 load from a 5-V power
supply. This and similar TI EVMs allow customers to evaluate the performance
of Tl's c1ass-O audio products without spending the time and resources normally
required to design and build a test circuit. In addition, each EVM is compatible
with the TI plug-n-play audio amplifier evaluation platform, which provides the
power, standard audio interconnects, signal conditioning, and speakers required
to operate the audio system.
TI class-O EVMs are available with or without an internal class-AS headphone
amplifier circuit. The ICs with the headphone circuit are equipped with the
necessary internal interface logic to select between the class-O and headphone
modes of operation. Each EVM includes onboard pushbutton switches for
manual muting and shutdown, and input pins for logic control of mode, mute, and
shutdown. A miniature stereo headphone jack is mounted on the EVMs that have
the internal headphone amplifier to allow convenient connection of headphones.
The modules have single in-line header connector pins mounted to the underside
of the boards. These pins allow the module to be plugged into the plug-n-play
platform, which automatically makes all of the signal input and output, power, and
control connections to the module. The module connection pins are on O.1-inch
centers to allow easy use with standard perf board- and plug board-based
prototyping systems, or for direct wiring into existing circuits and equipment when
used stand-alone.
These EVMs and the plug-n-play platform can be found at the TI web site:
http://www.ti.com/sc/apa.
TI and PowerPAD are trademarks of Texas Instruments Incorporated.

5-7

Class-O Amplifier Circuits

2 Class-D Amplifier Circuits
The class-O amplifier IG consists of an analog input circuit section, switching
circuit, a pulse-width modulation circuit, charge pump and gate drive circuit, and
an output circuit. All of these circuits, except the pulse-width modulator, require
external components for operation. This section focuses on the criteria for
determining these external components.

2.1

Input Circuit
The input stage of each channel of the class-D amplifier is a differential amplifier,
which means filters are required for both the noninverting and the inverting inputs
as shown in Figure 1. These input filters serve two purposes: they set the low
frequency corner, flO, and they block dc voltages and currents.

Voo

Class-O Amplifier IC

LlNP

Left
Channel
Inputs

LINN

10kQ

e----1f-----,
RINP

Right
Channel

Inputs

VOO

~W

RINN

10kQ
~

10kQ

VREF

Figure 1. Class-D Input Circuit and Filter
Each filter consists of one external capacitor (GIN) in series with the internal
resistance (RIN) of the amplifier input. Such a configuration creates a first-order,
high-pass filter (HPF) with a -3 dB cutoff frequency of
fLO

= _1_

2·1t·R·C
where R = RIN 10 kQ ±20% (typical) and G GIN 1 ~F for a -3 dB value of
15.~ Hz for the class-D EVMs. The fLO can be easily adjusted by changing the
value of CIN.
The inputs can also be driven single-ended by applying the audio input signal to
the non inverting input and ac-grounding the inverting input as shown by the
dashed line in Figure 1. This is necessary to avoid mismatching the impedance
of the two inputs, which creates a differential voltage and a potential for popping
in the speakers when power is applied to the system. The capacitor also prevents
dc current flow from the internal voltage reference to ground.

=

5-8

SLOA031

=

=

(1)

Class-D Amplifier Circuits

The internal gain of the class-O amplifier limits the input voltage to a maximum
of

v-~
IN Av

(2)

where Po is the maximum output power, RL is the dc load resistance, and Av is
the internal gain of the class-O amplifier. The large gain and low input currents
of the class-O amplifier reduces the input voltage to much less than 1 V and allows
the use of small, ceramic capacitors on the inputs.
The input capacitors should be placed as close to the input pins as possible to
reduce noise pickup. Connecting the inputs differentially further reduces the input
noise. Surface-mount, ceramic capacitors are readily available in 0805 for X7R
and Y5V, and can even be found in 0603 Y5V. Ceramic capacitors are preferred
over electrolytic for their small size, low equivalent series resistance (ESR), low
noise, and longer life of the component.

Design Considerations for Class-D Audio Power Amplifiers

5-9

Class-D Amplifier Circuits

2.2 Output Circuit
The class-D amplifier outputs are driven by heavy-duty DMOS transistors in an
H-bridge configuration. These transistors are either fully on or off, which reduces
the ROSON and the power dissipated in the device, increaSing efficiency. The
result is a square-wave output signal with a duty cycle that is proportional to the
amplitude of the audio signal. There are several options available as to what type
of filtering should be used to recover the audio signal. The output may be directly
applied to the speaker if the speaker is inductive at the class-D switching
frequency and EMI is not an issue, or a half filter could be used. 1 However, for
this application it is assumed that EMI is a consideration, and the focus is
therefore the full output filter shown in Figure 2.
RPVDD
Class-D
AmplHier IC

PVDD

GATE
DRIVE

PVDD

GATE
DRIVE

ROUTN

PVDD

GATE
DRIVE

PVDD

GATE
DRIVE

LPVDD

Figure 2. Class-O Output Circuit and Filter
5-10

SLOA031

Class-D Amplifier Circuits

The main goal of the output filter is attenuation of the high frequency switching
component of the class-O amplifier while preserving the signals in the audio band.
This describes the characteristic of a low-pass filter (LPF), which is specified by
its cutoff frequency (-3 dB point), gain and ripple in the pass band, and
attenuation in the stop band. The order of the filter determines how many poles
exist at the same frequency, with each order increasing the attenuation above the
cutoff frequency by -20 dB per decade. The switching frequency (fS) of the
class-O amplifier can influence the choice of the filter order - the higher the fS'
the lower the order required to achieve a given attenuation within a specified
passband. This would seem to dictate the use of the highest switching frequency
possible. The tradeoff is that increasing fS increases the switching losses and the
EMI, and decreases the efficiency of the amplifier.
A second order LPF reduces fS by -40 dB per decade to one percent of its
prefiltered value. A 5-V signal at 250 kHz is reduced by -40 dB over one decade
to 50 mV. If increased attenuation is desired, two alternatives remain; a higher
order filter could be implemented, increasing the number of components and the
cost, orfS could be increased, lowering the overall efficiency and increasing EMI.

2.2.1

Filter Design
The output filter is a simple, second-order, LC-type filter designed using a
Butterworth approximation. This type of filter is desired for the relatively flat'
pass-band response it provides and the small number of parts it requires. The
transfer function for a second order Butterworth approximation is
H(s) =

52 +

(3)

1

.f2 5 + 1

The first step is to realize the circuit and derive the transfer function, beginning
with a half circuit model and moving to the full-bridge circuit. The half circuit model
of the BTL output is shown in Figure 3, with half of the desired dc load resistance
(RH) of the speaker shown. The input signal (VIN) is the 250-kHz square wave
output of the class-O amplifier, while the output (VO) is the voltage developed
across the speaker.
Class-D

output

-+-i"---!~
VIN

-1

I

T

CH

i

RH

~o

Figure 3. BTL Half-Circuit Model

Design Considerations for Class-D Audio Power Amplifiers

5-11

C/sss-O Amplifier Circuits

Converting the inductance and capacitance into S-domain representations
( L=> Ls and C => 1/Cs), solving for the transfer function, and manipulating the
terms into the form of equation 3 gives the transfer function for the half-circuit
model.
1

H(s)

= Vds) =
V1N(S)

~

(4)

+ _1_ s + _1_

82

RH'CH

LH'CH

Equating the s terms and the real terms of equations 3 and 4 provide the
half-circuit values for CH and LH, respectively, These values are for the case
where mO 1 radian per second and should be frequency scaled by dividing
27tfC.
through by

=

roo =

CH

= __1_ =

L

=..1. = /2 . R = /2.

H

/2 ' RH
CH

1

(5)

2 . :It • fc . /2 . RH
2·

H

:It

(6)

RH

'fc

Two half-circuit models are then combined to yield the actual BTL circuit as shown
in Figure 4. The capacitors and resistors are then combined to provide the final
BTL equations.
L

L

Figure 4. Combination of Two Half-Circuit Models
(7)

RL = 2 . RH

CL =
L

1

2/2 . :It '

(8)
RL ' fc

= L = /2 ' RL = /2,
H

2.

00 0

4'

RL

:It •

(9)

fc

The inductor values actually remain the same for the half- and full-bridge circuit
since there are two inductors in the BTL circuit. The -3-dB cutoff frequency for
the LC filter, based on the BTL values, is
(10)

where the J2 in the denominator is the result of transposing the values for L
and C from the half-circuit model to the full BTL circuit.
5-12

SLOA031

Class-D Amplifier Circuits

Table 1 shows values for Land CL for a given fC and RL'

Table 1. Second-Order Butterworth LCL Values
DC LOAD
RESISTANCE (RL - 0)

CUTOFF FREQUENCY
(fC- kHz)

INDUCTOR VALUE
(L-llli)

CAPACITOR VALUE
(CL-IlF)

4

20

22.5

1.41

4

25

18

1.13

4

30

15

0.94

4

35

12.9

0.80

8

20

45

0.70

8

25

36

0.56

8

30

30

0.47

8

35

28

0.40

The capacitors labeled C in Figure 2 serve as high frequency bypass capacitors,
and are empirically chosen to be approximately 10% of 2 . CL. Their small value
has a negligible impact on the filter cutoff frequency.
The choice of filter components and fC may dictate the use of a series RC Zobel
network placed in parallel with the load. 1 This depends on the Q of the circuit,
which changes when a speaker, which is highly reactive, is connected as the load.

2.2.2 Design Example
The class-D audio system will have a passband of 20 Hz to 20 kHz and a
switching frequency (fS) of 250 kHz. The pass-band attenuation of fS should be
40 dB, and the corner frequency of the LPF will be set to avoid attenuating audio
signals by more than 1 dB across the audio spectrum. The speaker dc resistance
is 4 Q. A second-order LC filter is to be used.· What inductor and capacitor values
are required?
The inductance and capacitance are calculated using the BTL equations:
CL =

L =

2.

1

1t •

.f2. RL

4 .

1t •

.f2 . RL • fe

fe

=

4·

=

2.

.f2 . 40
1t .

1

1t •

25 kHz

.f2 . 40 . 25

kHz

=1.1IlF

= 18 H
Il

(11)
(12)

These values are checked by substituting into equation 10 and found to be
correct. Reviewing available component values shows options for L of 15 /J.H and
22 /J.H, and the closest value for CL is 1 /J.F. The values for CL =1 /J.F and L =15 /J.H
push the filter cutoff frequency out to 29 kHz.
The filter is now complete, except for the high frequency bypass capacitors
labeled C in Figure 2. These capacitors should be approximately 10% of 2 . CL,
or 0.2 /J.F. The nearest standard value of 0.22 /J.F is selected.
Design Considerations for Class-D Audio Power Amplifiers

5-13

C/ass-D Amplifier Circuits

OUTP

OUTN

-----,;:15~H h f. ~ L-rf1
r--LlJ
Figure 5. Complete BTL Output Filter

2.2.3 Component Selection
The output inductors are the key elements in the performance of the class-D
audio power amplifier system. The most important specifications for the inductor
are the dc resistance and the dc and peak current ratings. The dc resistance
directly impacts the efficiency by adding to the total load resistance seen by the
power supply. An approximation of the efficiency is
POUT

11 =

""""PIN =

12· RL
12 [2 (RosoN + RIND) + Rd

(13)

where RL is the dc resistance of the speaker, ROSON is the on resistance of the
DMOS power transistors, and RIND is the dc resistance of the inductors.
The inductor current ratings must be high enough to avoid magnetic saturation,
which will cause an increase in audio signal distortion or, if completely saturated,
will cause the inductor to appear as a short rather than an open circuit to the PWM
output. This could potentially damage the device or speakers from the resulting
high current surge that may occur during turn on, or the increased quiescent
current during normal operation. It would seem best, then, to choose an inductor
that has a much higher curren.t rating. The tradeoff is that the size and cost
increase as the current capabiiity increases. Shielded inductors will also help
reduce distortion and EMI, minimizing crosstalk in the process.
The filter capacitors should be ceramic capacitors with X7R characteristics for
stability over voltage and temperature, and can be found in common
surface-mount packages as small as 0805. The values of capacitance calculated
in the example above are readily available in ceramic chip and metal film
capacitor product lines. Measurements have shown little difference between the
performance of these two types of capacitors, though some audiophiles will
strongly recommend the metal film. The capacitors should be rated to handle the
sum of the dc and ac voltages, which will be

VCAP

=

(VS~PLY)

+ (0.707 . jP MAX

' RL)

where V SUPPLY is the power supply input voltage, PMAX is the maximum rms
power output for the amplifier, and RL is the dc resistance of the speaker. This is
the minimum supply voltage needed, and allowances must be made for
temperature, applied voltage, and transient voltage spikes. As a rule of thumb,
the voltage rating should be twice what is calculated.

5-14

SLOA031

(14)

Class-D Amplifier Circuits

2.3 Charge Pump Circuit
The charge pump circuit consists of one or more external charge pump
capacitors, an external charge storage capacitor, and an intemal circuit that
controls the flow of charge in the circuit. Figure 6 shows the internal and external
components and functions that make up a tripler charge pump circuit where CCP1
and CCP2 are the charge pump capacitors and CVCP is the charge storage
capacitor.
OMOS Gate 1--_ _C_I_as_s-_o_Am_p_lifi_'e_rI,C
Control

+
VCP ::r-CVCP
Inverter
Charge
Pump
Control

03

VCP2

CcP2

-=-

02
01
VSUPPLy-l~"""---+----'

VCP1

CCP1

VIN
Buffer

Figure 6. Tripier Charge Pump Circuit
VIN is a switching waveform thattransitions between VSUPPLY and 0 V. When VIN
is low, the output of the buffer is low, 01 is on, and CCP1 charges to VSUPPLY. The
inverter then provides a high output voltage to CCP2, 02 remains off, preventing
any charge transfer from CCP1 to CCP2, and 03 turns on. Charge is then shared
between CCP2 and CVCp. When VIN goes high the buffer output goes high, and
the voltage across CCP1 becomes (2 . VIN), turning 01 off. The inverter output
simultaneously provides a low output to CCP2, turning 02 on and 03 off. Charge
from CCP1 is then shared with CCP2. This process continues until the charge
builds up and VCP is in the operational range of (VSUPPLY + 6V) to (3 . VSUPPLY)
for a charge tripier, and (VSUPPLY + 6V) to (2 . VSUPPLY) for a charge doubler. The
charge from CVCP is then used to drive the OMOS output transistor gates.
The value for VCP must be large enough to supply the charge required by the
OMOS gate capacitance, yet small enough to fully charge within one-half of the
class-O switching period. Ifthese conditions are not met, CVCP fails to fully charge
during each switching cycle the ROS(ON) can increase substantially and degrade
the operation of the OMOS output transistors.

Design Considerations for C/ass-D Audio Power Amplifiers

5-15

Class-D Amplifier Circuits

The proper capacitance is recommended in the device data sheets and in
evaluation module user guides. The values required for these capacitors are
relatively small and are readily available in surface-mount ceramic chips. The
capacitors must be relatively stable over the expected operating temperature.
Good quality X7R, ±1 0% ceramic capacitors should be used with voltage ratings
greater than the maximum voltage of the charge pump, VCp, stated in the device
data sheets. Power dissipation is not a factor in this circuit as the currents are low
and the frequency of operation is high.

2.4 Switching Circuit
The switching circuit consists of a ramp generator and compensation capacitors
for each channel. These circuits all require external capacitors in order to
function. Selection of these capacitors is important for providing a balanced
triangular waveform and accurate regulation of the duty cycle for the output
transistors. The switching circuit is identical for each channel of the class-D
amplifier. Figure 7 shows the switching circuit for the right channel.

RPVDD
Charge Pump
Circuit

RPVDD
PVDD
VDD

RINP

ROUTP
ROUTP

RINN

101<.0

101<.0
VREF
ROUTN
ROUTN

Class-D Amplifier IC

COSCT

TCCOMP

Figure 7. Switching Circuit for the Right Channel

5-16

SLOA031

C/ass-D Amplifier Circuits

The ramp generator is the heart of the class-D amplifier - it sets the operational
frequency for the system from 100 kHz to 500 kHz. Oscillator capacitor COSC
charges and discharges at a constant rate with an applied constant current to
form a triangular waveform that is applied to one input of the comparator. The
capacitance is directly proportional to the period - doubling the capacitance
doubles the length of the period, decreasing the switching frequency (fS)' The
data sheets and EVM user guides provide the value of capacitance required to
generate a nominal fS of 250 kHz. Knowing the value of this capacitance (C250),
fS, and the desired switching frequency, the new capacitance, C, can be easily
calculated for any desired frequency of oscillation, f, from the ratio of two
capacitors as shown in equation 15.
Cose = C250

•

(!r)

(15)

The compensation capacitors, CCOMP are used to stabilize the comparator inputs
and should be identical to COSC' Ceramic capacitors with COG temperature
characteristics are the common type available in such a small capacitance.
These capacitors do not exhibit a change in value with changing ac or dc
voltages, and are extremely stable over large temperature ranges. A standard
50-V COG-type capacitor with a maximum of ±5% tolerance is recommended,
with much tighter tolerances available if desired.

Design Considerations for C/ass-D Audio Power Amplifiers

5-17

Headphone Circuit

3 Headphone Circuit
Some of the class-D amplifier ICs feature class-AB headphone (HP) amplifier
circuits capable of driving 50 mW of power into a 32-0 load from a 5-V supply.
TTL-compatible interface logic (a mode pin) is provided to select between class-D
or class-AB modes of operation. Each HP channel consists of an internal
operational amplifier and pins for connecting external components that control
the gain and filtering for the headphones.
Class-D EVMs are available that integrate the HP amplifier functions. A typical
channel of the HP circuit for such an EVM is shown in Figure 8. External pins on
the EVMs allow easy connections to the inputs and outputs, and a miniature
headphone jack has been provided on the EVM board for easy testing of the HP
amplifier. The HP jack includes the control pins necessary to control the IC mode.
An onchip regulator provides the 5 V required for operation of the HP amplifier
circuit. The power decoupling capacitor, C, is discussed in the Device Power
Supply Decoupling section of this report. Capacitor CV2P5 stabilizes the HP
circuit, and should be the size recommended in the data sheets and the EVM user
guides.

HPDR

PVDD

V2P5

HPRIN

From Left
Channel
CIN

T

Audio Input

VSUPPLY

Figure 8. Headphone Amplifier Circuit, Right Channel
Each amplifier is configured as an inverting operational amplifier with externally
controlled gain. The transfer function for this circuit,. ignoring COUT, R, and any
load resistance, RL, is shown in equation 16 where 0)1
(CF . RF)-1 and
0)2 =(CIN . RIN)-1.

=

HOw) = Vo =

(-

;~;)

V (1 + ~) (1 + ~)
IN

5-18

SLC>A031

(16)

Headphone Circuit

Input capacitor CIN serves to ac-couple the input. The series combination of RIN
and CIN in this circuit creates a LPF function in the denominator, which then acts
as a HPF to set the low frequency corner shown in equation 17, where R RIN
and C CIN. fLO can be easily adjusted by changing CIN or RIN.

=

=

fLO = _1_
2·:n;·R·C

(17)

Capacitor CF is recommended for stability purposes when the gain is greater than
or equal to -10 VN. The parallel combination of RF and CF then creates a HPF
function which, when in the denominator, acts as a LPF to set the high frequency
corner (fHI) of the circuit. equation 17 may be used to calculate fHI, with R RF
and C CF. This corner frequency should be about 300 kHz, well above the audio
band.

-=

=

Capacitor COUT is required for all single-ended audio circuits to ac-couple the
output, preventing dc current from flowing into the HP. COUT forms another LPF
in conjunction with the dc resistance (Rdof the headphones. Resistor R may be
included if the IC mode control interface is implemented with the HP jack, and is
much larger than RL and can be ignored in this analysis. The class-O EVMs with
HP amplifiers use such a circuit. equation 17 is again used to calculate the low
frequency corner for this filter. It should be noted that the corner frequencies of
the input and output filters will overlap to some degree.
The HP circuit includes some internal depop circuitry that is used to minimize the
pop in the speakers when the HP is activated and deactivated. The largest
capacitor that is recommended for use with this circuit is 331lF. Higher values may
be used, but will decrease the effectiveness of the depop circuit.
Ceramic capacitors are available for the small values of capacitance used for the
input and feedback path. The voltage rating of the input capacitor will depend
upon the gain of the circuit, which should be greater than the passband gain (AV)
in equation 18.

Av =

IRFI
RIN

(18)

This is then used to calculate the maximum input voltage in equation 19.
(19)

VIN = 5V

Av

The voltage rating of the feedback capacitor should be a minimum of 5 V, and is
readily available in a ±5% COG package for such a low capacitance. The input
capacitors are larger and available in a ±1 0%, X7R package, depending upon the
value.

DeSign Considerations for Class-D Audio Power Amplifiers

5-19

Control and Indicatgr Circuits

4 Control and Indicator Circuits
The Texas Instruments class-O audio power amplifiers have three main control
input pins (shutdown, mute, and mode) for external control of chip functions. Each
of these inputs is TTL compatible to allow easy interface with logic. The shutdown
and mute controls are provided with each class-O device, whife the mute control
is only applicable to devices that incorporate a class-AB headphone amplifier.
Two indicator pins (faultO and fault1) are also provided to allow monitoring of chip
status. They provide feedback when an under-voltage, over-current, or thermal
fault exists. These pins are provided on each of the devices.

4.1

Shutdown
The shutdown control pin allows the device to be placed into a power-saving
sleep mode to minimize current consumption. This pin is TTL active low - a
voltage of less than 0.8 V at this pin will shut down the entire device. The device
will become active when the voltage at the pin rises above 2 V. When in shutdown,
the Ie draws a maximum quiescent current that is less than 1 !JA.
In typical applications, as often found in, notebook computers, portable audio
products, and such, the internal speakers mute when headphones are plugged
into the headphone jack, or internal speakers mute when external speakers are
connected. In applications using separate speaker and headphone amplifiers,
the one not being used can be shut down to conserve power.

4.2 Mute
The mute control pin turns on the low-side output transistors, shorting the load
to ground and muting the outputs of the device. This pin is TTL active low - a
voltage of less than 0.8 V will mute the device outputs. The outputs will tum on
when the voltage at the mute pin rises above 2 V. When muted, the class-O device
draws only a few mA of quiescent current.

4.3

Mode
The mode control pin selects either the class-O or the headphone amplifier as the
active amplifier, placing the inactive amplifier in a power-saving sleep mode. This
pin is TTL compatible, with a voltage less than 0.8 V activating the class-O
amplifier, and a voltage greater than 2 V activating the headphone amplifier.

5-20

SLOA031

Control and Indicator Circuits

This function can easily be controlled with a headphone jack that contains an
internal switch to change the state of the control line, and has been successfully
implemented on the EVMs for the class-D amplifiers that integrate headphone
circuits. Figure 9 shows an example of this type of circuit.
Class-D
AmplifierlC

VSUPPLY
R1

MODE
HPDR

1

If

HPROUT

1\

3

C

R2
If

HPLOUT

I~

HPDL
~

Ir

To
To
FEEDBACK FEEDBACK

1

B

t

~

-==-

Headphone
Jack

2

R3

~

1

1

Figure 9. Mode Control Circuit Featuring Headphone Jack Control
Resistors R1 and R2 form a divider network when a headphone plug is not
inserted into the headphone jack. The ratio of these resistors should be such that
the mode pin is held below 0.8 V to activate the class-D amplifier. When a
headphone plug is inserted into the jack, contact B is disconnected from pin 3 of
the jack and no current flows through R 1, causing the mode pin to float to V SUPPLY.
This deactivates the class-D amplifier and activates the headphone amplifier.
Removal of the headphone plug from the jack then connects contact B to pin 3
and pulls the MODE pin low, causing the device to revert to class-D operation.
Resistor R3 is included in the remaining channel to balance the outputs of the two
channels when the headphone amplifier is active.

4.4

Fault Indicators
Two fault indicator pins on the class-D amplifier Ie provide feedback when a fault
condition exists. Signals on these pins indicate the status of the class-D amplifier:
operational, over-current, thermal fault, and under-voltage lockout. The only
status reported for the class-AB headphone amplifier is for a thermal fault, which
is indicated by the same error code as for the class-D amplifier. The device data
sheets list the error codes for each of these conditions.
The TTL-compatible fault pins are connected to open drain outputs and require
a pullup resistor to limit the current flow into the pins to a maximum of 1 mA. Once
a fault is triggered, the appropriate fault pins remain active until the fault is cleared
by cycling the shutdown pin, mute pin, or the power supply to the device.

Design Considerations for Class-D Audio Power Amplifiers

5-21

Device Power Supply Decoupling

5

Device Power Supply Decoupling
Adequate delivery of power and proper grounding reduces distortion and ensures
correct operation of the class-D device. Power supply filtering and appropriate
ground connections are discussed below.
Power supply filtering has two objectives: decouple the power supply from the
class-D amplifier and provide a path for high frequency noise to bypass the
device. There are three main power inputs for the device: class-D analog input
and controls (VDD), charge pump and headphone (PVDD), and the output
(RPVDD and LPVDD). Figure 10 shows the power bus and recommended
filtering for a class-D audio power amplifier.

VSUP PLY

-;: :::::

-;:![:;:~

VDD

0

~

0

I I I

Charge
Pump
Circuit

Analog

AGND

PGND

i

PVDD

~

Headphone
& Charge
Pump

I

I;; :==:;::

LPVDD

r'CPVDQ

RPVDD

Class-D Power Output
Stage

;; ~
'CLPVDD

v SUPPLY

r:c

B1

I

PGND

AGND

;;.. ~

;;: ::::""CVDD

CB2

CB2

CB1

-::
CRPVDD

r:

GND

GND

Figure 10. Class-D Power Bus, 48-Pln TSSOP Package
All of the capacitors connected to the power bus (VSUPPLY) are working to
decouple the circuit from the power supply. The large bulk capacitors (CS1 and
CS2) are provided for each channel to supply the majority of the switching current
required by the amplifier. Smaller capacitors (CVDD, CPVDD, CLPVDD, and
CRPVDD) are placed adjacentto the various power pins to supply the initial charge
of the switching current. The only power pins located on the right side of the chip
(RPVDD) are for the high power output section of the right channel. The
remaining power pins (VDD, PVDD and LPVDD) are located on the left side of
the chip and will be the focus of the discussion. The right channel capacitors will
then be identical to those of the left channel.

5.1

Bulk Capacitors
Real-world capacitors are modeled using parameters such as equivalent series
resistance (ESR), equivalent series inductance (ESL), capacitive reactance (XC)
and inductive reactance (XL>. The equivalent impedance of a capacitor over
frequency is simply modeled by

Z = jESR2 + (Xc - Xd 2
5-22

SLOA031

(20)

Device Power Supply Decoupling

XL is small for frequencies below 1 MHz and can be neglected since the switching
frequency range of the TPA005D14 is 100 kHz to 500 kHz. The capacitive
reactance is maximum and dominates at dc. It decreases as the frequency
increases until resonance is reached (XC = XL>, at which point Z = E5A. The E5R
of a capacitor is considered to be constant over the 100 kHz to 500 kHz switching
frequency range of the class-D amplifier, and is usually provided by the
manufacturer.
The values for the bulk capacitors CS1 and CS2 are the primary concern, and are
calculated using the circuit shown in Figure 11. It is assumed that LIN is large
(steady current flows from the power supply) and has a negligible ripple, the
capacitor current for C is negligible, and the switching frequency and dc load
resistance is known.

'-IN

VDD

+

Co

Power Supply and Filter

I
I

Figure 11. Power Supply Bulk Oecoupling CapaCitor Circuit
The peak power for a given load is then used to calculate the peak voltage, which
is then used to calculate the peak current.
=

V PEAK

IPEAK --

jPPEAK •

RL

(21)
(22)

V PEAK
R;:-

This current flows from Cs through 51, the load, and 52 to ground. The minimum
capacitance required to supply this peak switching current is

c=

IpEAK ' To' DMAJ(

(23)

VRIPPLE

where T D = 1/fswitch is the period, DMax is the maximum duty cycle, and VRipple
is the desired ripple voltage, or droop, that will appear at the output of the
amplifier. This is the capacitance required to limit the ripple voltage based on the
capacitance alone. In most every case, the ripple voltage caused by the E5R will
dominate. The maximum E5R required to achieve the same VRipple for the same
IPeak from equation 22 is calculated in equation 24 below.
ESR

= VRIPPLE

(24)

IPEAK
Design Considerations for Class-D Audio Power Amplifiers

5-23

Device Power Supply Decoupling

The total ripple voltage contributed by the bulk capacitor Cs is the sum of
equations 23 and 24. The requirements of the application will determine the
acceptable tradeoffs in the selection of components that meet these criteria. It
should be noted that the total ripple voltage seen at the output of the class-D
amplifier wil.1 be approximately equal to that calculated in equation 25.

V~PLE = IPEAK[ (To ' gMAX) + ESR + ROS(ON)]
There are various ways to implement the bulk capacitance that is selected: one
large capacitor that meets the requirements of both equations (23) and (24) can
be used; two or more capacitors can be paralleled to reduce the ESR and the size
of the capacitors; or two different types of capacitors can be used to supply the
current and meet the ESR specifications. Keep in mind that the ESR of the actual
capacitor used should be 30% - 50% lower than the calculated value to allow for
increases due to temperature, ESL, and aging.
Electrolytic capacitors, aluminum or tantalum, are the best choice right now for
large capacitance requirements, though ceramic capacitors of up to 100 IJ,F are
being produced in low voltage packages. The electrolytic capacitors are normally
useful for applications below 1 MHz. This is due to their low resonant frequency
and is the reason for using smaller, ceramic capacitors in parallel with the
electrolytic. Electrolytic capacitors, in particular the tantalum type, are subject to
damage by stress from exceeding the voltage rating. They must be chosen such
that they will retain the minimum required capacitance and maximum ESR over
the entire temperature range and for the voltage range to avoid damage and early
failure of the components. The voltage rating should be greater than the sum of
the supply voltage and the total maximum ripple voltage of equation 24.

5.2

Small Oecoupling Capacitors
The large capacitance of CS1 and CS2 means a slower response time due to the
large time constant formed with the resistance of the circuit, and is why the
smaller capacitors CVDD, CPVDD, CLPVDD and CRPVDD are used. These
capacitors provide a smaller time constant for a much quicker discharge time, and
supply the initial transient charge required for the high frequency switching pulses
of the class-D amplifier. Their low value pushes the resonant frequency of the
capacitor out - they appear capacitive at much higher frequencies due to the
smaller XL of equation 20. This serves to bypass unwanted high frequency
signals.
The current for the VDD pin is very low and can have the transient requirements
satisfied by a 0.1 IJ,F or 1 IJ,F capacitor. The PVDD pin will draw less than 100 mA
of current and should have a 1-IJ,F decoupling capacitor. These must have a
voltage rating that is greater than the sum of the supply voltage and the maximum
ripple voltage of equation 25.
The values required for these capacitors are small and readily available in
surface-mount ceramic chips. The capacitors should be relatively stable over the
expected operating temperature. Good quality X7R, ±10% ceramic capacitors
are available for the capacitance required, though ±20% or +80/-20% Y5V
capacitors may be used, depending upon the application. Power dissipation is a
factor in this circuit as the currents can be quite high.

5-24

SLOA031

(25)

PCB Layout

6 PCB layout
Good layout practices and well thought out design provide excellent performance
for the TI class-D audio power amplifiers. There are three main areas of concern
in the layout: the ground plane, power plane, the inputs and the outputs. Each is
discussed briefly below. See the TI website for more information on class-D
layouts.

6.1

Ground Plane
Experimentation with several types of ground planes has shown that, with some
careful planning and good layout practices, a solid ground plane works as well
as other types of grounding schemes. This is due in part to the relatively low
frequencies of operation for the system, and to the careful layout of the
components and traces. The solid ground plane also serves to assist the
PowerPAD2 in the dissipation of heat, keeping the class-D amplifier relatively cool
and negating the need for an external heat sink. Connection to the PowerPAD is
discussed later in this section. Finally, the ground plane can act as a shield to help
isolate the power pins from the output, reducing the impact of EMI on the traces
and pins.
It is important that any components connecting an IC pin to the ground plane be
connected to the nearest ground for that particular pin. Table 2 lists the ground
pins for the various sub-circuits that are part of the TPA005D14 class-D IC to
assist in determining where a component should be grounded. Care should be
taken to prevent the ground return path of any high current components (such as
the output filter capacitors) from directly passing through other ground
connections of the IC, particularly the input.

Table 2. Audio Power Amplifier Subcircuit Ground Pins
GROUND PIN No.

47

12,13,36,37

TPAOO5D14 RELATED PINSt

APPLICABLE CIRCUITS
Controls (shutdown, mute, mode)

1,2,3

Class-O outputs

4,5,44,45

Ramp generator

6,43,48

Grounds

7,46

Input power (VDO)

8

Fault Indicators

41,42

Output power (LPVOO, RPVOO)

9,16,33,40

Class-O outputs

10, II, 14, 15,34,35,38,39

20

Headphone

17,18,19,23,26,29,30,31,32

27

Charge pump

21,22,23,24,25,26,28

t Pin numbers may vary in other class-O devices.

Design Considerations for Class-D Audio Power Amplifiers

5-25

PCB Layout

6.2

Power Plane
There are three main power sections on the chip: the input circuit power pins
(VDD), the output stage power pins (LPVDD and RPVDD), and the power for the
headphone and charge pump circuits (PVDD), as shown in Figure 9. When the
device is operating (Le. audio is being applied to the amplifier), the VDD pin draws
only a few mA of current and the PVDD pins draw several tens of mAo This is in
sharp contrast to the amps of current drawn by the LPVDD and RPVDD pins.
The power traces are kept short and the decoupling capacitors placed as close
to the power pins as possible. This is particularly true for the small decoupling
capacitors that are to be placed adjacent to each Ie power pin. Terminate
the capacitor ground close to the ground for the particular power section as
possible while paying attention to ground return current paths. This minimizes
ground loops and provides very short ground return paths and high frequency
loops.
The VDD pin supplies power for sensitive analog circuitry and is the most
sensitive pin of the device. It must, therefore, be kept as noise free as possible.
The demand for peak current is small and mostly satisfied by the charge of the
small decoupling capacitor. The PVDD pin(s) are not as sensitive to noise as the
VDD pin. They supply the current for the headphone regulator and control circuits
when the device is in class-AS mode (when applicable), and the charge pump
circuit when in class-D mode. The power traces for these power inputs should be
connected to the main power bus at a point near the large decoupling
capacitor(s). The small inductance of the traces and the charge supplied by the
large decoupling capacitor greatly reduces the ripple current of the main power
bus seen by these pins. Terminate the capacitor ground side close to the ground
for the particular power section while paying attention to ground return current
paths. Again, this minimizes ground loops and provides very short ground return
paths and high frequency loops.
The main power bus should terminate into the LPVDD and RPVDD pins, with the
small decoupling capacitors for each channel placed adjacent to each pair of
pins. When more than one bulk capacitor is used, place the smaller of the two
between the power pins and the large bulk capacitor. These traces should be
wide enough to handle the maximum peak current per channel over the operating
temperature range, and symmetric to facilitate even power distribution. Place
them directly over the ground plane to reduce EMI and minimize the ground return
path.

5-26

SLOA031

PCB Layout

6.3

Inputs and Outputs
The pinout of the class-D amplifiers facilitates the separation of the inputs and
outputs, enabling isolation of ground return paths and high frequency loops. The
class-D and headphone amplifier input traces should be kept as short as possible
between the ac coupling capacitors and the amplifier IC input pins to reduce noise
pickup. Keep the inputs separated from the outputs, particularly from the
inductors if unshielded units are used, to minimize magnetic coupling. The
headphone traces may be in close proximity with the class-D output since the two
amplifiers are not active at the same time.
The control (shutdown, mute, and mode) input pins have almost no current flow
through them, and inductance and resistance of the traces is of a minimal
concern. The indicator output pins (faulW and fault1) have less the 1 mA of current
flow, and should be sized accordingly. There are no special considerations for the
layout of these traces - standard layout practices will apply.
It is critical to minimize the trace lengths between the device class D output pins
and the LC filter components, particularly those that contain the full square wave.
The traces to the inductors should be kept short, yet separated from the input
circuit as much as possible. Routing the pre-inductor output traces of a particular
channel (Le., ROUTP and ROUTN) on adjacent layers so that they overlap will
cause the magnetic fields to subtract from each other, reducing the EM!. All
high-current output traces should be wide enough to allow the maximum peak
current to flow over the entire operating temperature range of the system. Failure
to do so will create excessive voltage drops, a decrease in efficiency, and an
increase in distortion.

6.4

General PowerPAD Considerations
The class-D IC is mounted in a special package that incorporates a thermal pad
designed to transfer heat from the silicon die of the IC directly to the PCB. The
PowerPAD'" package is constructed using a downset leadframe. The die is
mounted on the leadframe with the chip ground tied to the pad through a low
impedance. The bottom surface of the leadframe is exposed and serves as a
metal thermal pad on the underside of the IC package. This metal is then soldered
directly to the PCB, providing direct contact between the die and the PCB etch,
which, in tum, provides an exceptional thermal transmission path. Excellent
thermal performance can then be achieved by providing this thermal path on the
PCB.
The following steps illustrate the recommended approach to properly heatsink a
TI class-D audio power amplifier 4S-pin DCA package that integrates the
PowerPAD with a circuit board.

Design Considerations for Class-D Audio Power Amplifiers

5-27

PCB Layout

1. Prepare the PCB for proper connection to the class-D IC with a top layer etch
pattern as shown in Figure 12. Etch should be provided for both the IC leads
and the PowerPAD.

111111111111111111111111
Thermal pad area (125 mils x 250 mils) with 21 vias
(Via diameter", 13 mils)

111111111111111111111111
Figure 12. PowerPAD PCB Etch and Via Pattern

2. Place 21 vias evenly spaced in three rows (seven per row) in the area for the
PowerPAD. These vias should be 13 mils in diameter to minimize solder
wicking through the holes during reflow soldering, ensuring a good
.
connection between the IC thermal pad and the PCB etch.
3. Additional vias may be placed anywhere along the thermal plane outside of
the PowerPAD area to assist with heat dissipation. These vias are not
restricted to the 13 mils of step 2 since they are not used to connect the IC
to the PCB.
4. Connect all of these vias to the PCB ground plane. The ground plane now
becomes the heatsink for the amplifier IC.
5. Do not use a web or spoke connection when connecting these vias to the
ground plane. Web connections have a high thermal resistance that is used
to slow heat transfer to the ground plane, making soldering of these vias
easier. This would impair the flow of heat between the PowerPAD and the
circuit board ground plane and is not recommended.
6. The solder mask on the top layer should then leave the etch pads for the IC
pins and PowerPAD exposed. The bottom layer solder mask should,
however, cover the entire thermal pad as well as the via edges, leaving tiny
holes in the very center of each via. This prevents the solder connecting the
IC thermal pad to the PCB from being wicked away during reflow.
7. Apply solder paste to the exposed etch pads for the IC pins and PowerPAD.
8. The class-D IC is then soldered in position during the reflow process. Actual
thermal performance achieved with the package will depend upon the
application. The Texas Instruments Technical Brief, PowerPAD Thermally
Enhance Package, Literature Number SLMA002, contains more information
on the PowerPAD package and its thermal characteristics.
5-28

SLOA031

References

7 References
[1] The Texas Instruments application report, Reducing and Eliminating the C/ass-D
Output Filter, literature number SLOA023.
[2] The Texas Instruments technical brief, PowerPAD Thermally Enhanced Package,
literature number SLMA002, contains more information on the PowerPAD package
and its thermal characteristics.

Design Considerations for Class-D Audio Power Amplifiers

5-29

5-30

SLOA031

Mono Configuration of the
TPA005D02 Class-D Audio Power
Amplifier
Application Report

Literature Number: SLOA028
July 1999

:'I
TEXAS
INSTRUMENTS

Printed on Racyclad Paper

5-31

IMPORTANT NOTICE

Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products orto discontinue
any product or service without nolice, and advise customers to obtain the latest version of relevant information
to verify, before placing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those
pertaining to warranty, patent infringement, and limitation of liability.
TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with Tl's standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of aU parameters of each device is not necessarily
performed, except those mandated by govemment requirements.
CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
DEATH, PERSONAL INJURY, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE ("CRITICAL
APPLICATIONS'~. TI SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, AUTHORIZED, OR
WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT DEVICES OR SYSTEMS OR OTHER
CRITICAL APPLICATIONS. INCLUSION OFTI PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO
BE FULLY AT THE CUSTOMER'S RISK.
In order to minimize risks associated with the customer's applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
Intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. Tl's publication of information regarding any third
party's products or services does not constitute Tl's approval, warranty or endorsement thereof.

Copyright © 1999, Texas Instruments Incorporated

Contents
Design Problem •..•••..•..••••.••..•.•.••..•.•.••...•..••••••.••.•••••.••••••..•........•..•... 5-35
Solution .••••••.•••.•••••••.••••.•••.••.•••...•........••••••.••••..•..••.•..•.......•..••.•••. 5-35
Conclusion •••...••.••••••••••••••••.•••.•••..••.......••••.•.••••.•••.•••••.••••.••.•••••••.•• 5-37

List of Figures
TPA005D02 Class-D EVM Schematic Diagram for Mono Configuration .............................. 5-34

Mono Configuration of the TPAOO5D02 Class-D Audio Power Amplifier

5-34

SLOA028

Mono Configuration of the TPA005D02 C/ass-D Audio Power
Amplifier
Edward A. Thomas
ABSTRACT
Class-D Audio Power Amplifiers (APAs) are becoming an extremely popular choice for
audio solutions in battery-powered applications. The increased efficiency and reduction
in heat dissipation of a Class-D APA versus that of a Class-AB APA allows the battery life
on an application to be extended. The TPA005D02 is monolithic stereo Class-D APA
offered from Texas Instruments. This document discusses how to configure the
TPA005D02 to be used in a mono configuration. The actual specifications of the
TPA005D02 can be found in the published Texas Instruments data sheet (literature
#SLOS227A).

Design Problem
Many battery-powered applications would like to take advantage of the increased
efficiency of the TPA005D02 APA but do not need stereo output. This document
will show the specific application circuit in a mono configuration. The use of this
device in the mono configuration saves board space, cost, and supply current
when compared with the same device used in a stereo configuration.

Solution
The use of the TPA005D02 APA in the mono configuration eliminates the need
for many of the surrounding components required to operate the device in the
stereo configuration. The schematic for the TPA005D02 APA is in the
TPAOO5D02 Evaluation Module User's Guide (literature #SLOU032A). The
modifications needed to be made to the evaluation board for the mono
configuration of the TPA005D02 are shown in the schematic shown in Figure 1.
The TPA005D02 APA integrated circuit consists of two separate amplifiers inside
the device, one for the right channel and one for the left channel. To operate in
the mono configuration, only one of the two amplifiers inside the TPA005D02 will
be used. The TPA005D02 has two pins (LCOMP and RCOMP) that can be used
to shut down power to the respective amplifier. Tying the respective xCOMP to
GND will stop the bridge from switching and will save quiescent power of the
device. In this document, the left amplifier will be shut down to allow operation of
the device in the mono configuration. In order to shut down the left amplifier,
LCOMP (pin 43) and input pins L1NP (pin 5) and LINN (pin 4), will be tied directly
to GND (see Figure 1). The operation of this device in the mono configuration
eliminates ten external components when compared with use of this device in the
stereo configuration. The capacitors on the inputs of the unused amplifier and on
the xCOMP will be eliminated from use in the mono configuration. The two
inductors and three capacitors on the output of the unused amplifier will also be
eliminated.
5-35

The Voo power supply pin sets for both amplifiers in the TPAOO5D02 must be
connected even though one amplifier (left in this example) is shut down. No power
will be pulled by the unused amplifier. The Voo supply pin sets are connected
through a guard ring internally, the device can be destroyed if only one supply pin
set is connected. The unused amplifier (see Figure 1) will not pull large current
transients through the power pins, therefore the 1 J.1F bypass capaCitor (C13) on
the LPVoo (pin 16) can be replaced with a 0.1 J.1F ceramic capacitor (shown). The
bypass capaCitors C15 (220 J.1F) and C11 (10 J.1F) on the unused channel may be
removed. The output pins LOUTP (10, 11) and LOUTN (14, 15), for the unused
amplifier, will be left floating.
The MUTE and FAULT features of the TPA005D02 Viill operate normally in this
mono configuration. The two detectable fault conditions are the charge pump
under-voltage lock-out condition and the thermal fault condition. More details on
the functionality of these features can be found in the product's data sheet.
Voo -~---<_-~---<_-. Voo
Voo

Voo

Conclusion
The Class-D APA is an effective, highly efficient, audio solution for many
battery-powered applications. A comparison of class D amplifier versus linear
amplifier supply current is included in the TPA005D02 datasheet. The results at
normal listening levels show the linear amplifier to have three times the current
draw of the class D device. This comparison is important in showing the selection
of the type of audio amplifier used in a battery-powered system can extend
battery life by three times, if a class-D amplifier is used. Offering flexibility in the
way to configure the TPA005D02 allows both mono and stereo configurations the
advantage of this increased efficiency in battery-powered systems. This allows
use of this device in many different applications that could benefit from Texas
Instruments, Class-D technology.

Mono Configuration of the TPAOO5D02 C/ass-D Audio Power Amplifier

5-37

5-38

PowerPADTM Thermally Enhanced Package

PowerPAD Thermally
Enhanced Package
TECHNICAL BRIEF: SLMA002

Mixed Signal Products

Semiconductor Group
21 November 1997

•
TEXAS
INSTRUMENTS

~TEXAS

INSTRUMENTS
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5-39

PowerPADTM Thermally Enhanced Package

IMPORTANT NOTICE
Texas Instruments (TI) reserves the right to make changes to ~s products or to discontinue any semiconductor
product or service w~hout notice, and advises its customers to obtain the latest version of relevant Information
to verify, before placing orders, that the information being relied on Is currant.
n warrants performance of Its semiconductor products and related software to the specifications applicable at
the time of sale In accordance ~h 11's standard warranty. Testing and other quality control techniques are
utilized to the extent n deems necessa/}' to support this warranty. Specific testing of all perameters of each
device Is not necessarily pertormed, except those mandated by government requirements.
Certain application using semiconductor products may Involve potential risks of death, personal injury, or
severe property or environmental damage iCr~lcal Applications").
n SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, INTENDED, AUTHORIZED, OR WARRANTED
TO BE SUITABLE FOR USE IN LIFE-SUPPORT APPLICATIONS, DEVICES OR SYSTEMS OR OTHER
CRITICAL APPLICAnONS.
Inclusion of TI products In such applications is understood to be fully at the risk of the customer. Use of n
products in such applications requires the written approval of an appropriate TI officer. Questions concerning
potential risk applications should be directed to TI through a leeel SC sales ofIIce.
In order to minimize risks associated with the customer's applications, adequate design and operating
safeguards should be provided by the customer to minimize inherent or procedurai hazards.
11 assumes no liability for applications assistance, customer product design, software performance, or
Infringement of patents or services described herein. Nor does n warrant or represent that any license, either
express or implied, is granted undar any patent right, copyright, mask work right, or other Intellectual property
right of n covering or relating to any combination, machine, or process In which such semiconductor products

or services might be or are used.

Copyright © 1997, Texas Instruments Incorporated

~TEXAS

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5-40

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PowerPADTM Thermally Enhanced Package

TRADEMARKS
nand PowerPAD are trademarks of Texas Instruments Incorporated.
MQUAD is a registered trademark of Olin Corporation
Other brands and names are the property of their respective owners.

~TEXAS

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5-41

PowerPADTM Thermally Enhanced Package

~1ExAs

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PowerPADTM Thermally Enhanced Package

Contents
Abstract......................................................................................................................... 7
1. Introduction ............................................................................................................ 8
2. Installation and Use ...............................................................................................10
2.1 PCB Attachment ..............................................................................................10
2.2 PCB Design Considerations ............................................................................ 11
2.3 Thermal Lands ................................................................................................12
2.4 Thermal Vias ...................................................................................................15
2.5 Solder Stencil Determination ........................................................................... 18
3. Assembly ...............................................................................................................20
3.1 Solder Reflow Profile Suggestion ....................................................................24
3.2 Installation and Assembly Summary ................................................................25
4. Rspair .....................................................................................................................26
4.1 Part Removal From PCBs ...............................................................................27
4.2 Attachment of a Replacement Component to the PCB ....................................28
5. Summary ................................................................................................................30
Appendix A. Thermal Modeling of PowerPAD Packages.........................................31
General ...................................................................................................................32
Modeling Considerations .........................................................................................32
Texas Instruments Recommended Board for PowerPAD ........................................33
JEDEC Low Effective Thermal Conductivity Board (Low·K) .....................................34
Boundary Conditions ...............................................................................................37
Results ....................................................................................................................38
Conclusions .............................................................................................................39
Appendix B. Rework Process for Heat Sink TQFP and TSSOP PowerPAD
Packages - from Air-Vac Engineering ........................................................................40
Introduction ..............................................................................................................40
Equipment .................................................................................;.............................40
Profile ......................................................................................................................42
Removal ..................................................................................................................42
Site Redress ............................................................................................................43
Alignment ................................................................................................................43
Replacement ...........................................................................................................44
Cone/uslon ...............................................................................................................44
Appendix C. PowerPAD Process Rework Application Note from Metcal ..............A5
Removal ..................................................................................................................45
Conduction Procedure .............................................................................................45
Convection Procedure .............................................................................................45
Placement Procedure ..............................................................................................46

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5-43

PowerPADTM Thermally Enhanced Package

Figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.

Schematic Representation of the PowerPAD Package Components .................. 8
Bottom and Top View of the 20 pin TSSOP Power PAD Package ..................... 10
64 Pin, 14x 14x 1.0mm BodyTQFP PowerPAD Package ............................... ll
Package and PCB Land Configuration for a Single Layer PCB ......................... 12
Package and PCB Land Configuration for a Multi-Layer PCB ........................... 13
64 pin TQFP Package with PowerPAD Implemented, Bottom View .................. 14
PCB Thermal Land Design Considerations for Thermally Enhanced TQFP
Packages .................................................................................................14
Figure 8. Impact of the Number of Thermal Vias versus Chip Area (Ole Area) ................. 16
Figure 9. Impact of the Number of 0.33mm (0.013 inch) Diameter Thermal Vias versus
Chip Area (Die Area) ................................................................................ 16
Figure 10. Ideal Thermal Land Size and Thermal Via Patterns for PowerPAD ................. 17
Using 100 pin PowerPAD TQFP
Figure 11. Test Board for Measurement of Sjo and
Packages .................................................................................................21
Figure 12. Typical Infrared Oven Proflle ...........................................................................25
Figure 13. Texas Instruments Recommended Board (Side View) ..................................... 34
Figure 14. Thermal Pad and Laad Attachment to a PCB Using the PowerPAD Package.35
Figure 15. General Laadframe Drawing Configuration .....................................................36
Figure 16. PowerPAD 8JC Measurement ...........................................................................37
Figure 17. Standard Package8Jc Measurement ............................................................... 38
Figure 18. Comparison of 8JA for Various Packages ........................................................39
Figure 19. DRS22C Reworking Station ............................................................................ 40
Figure 20. Reworking Nozzles of Various Sizes ...............................................................41
Figure 21. Nozzle Conflguration .......................................................................................42
Figure 22. Alr-Vac Vision System ....................................................................................43

e..

Tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.

Typical Power Handling Capabilities of PowerPAD Packages .............................. 9
Measured Sjo from Test Board ...........................................................................22
Measured 9ja from Test Board ...........................................................................22
Relationship of the Solder Joint Area on SIc, from Test Board Data ................... 23
Relationship of the Solder Joint Area on aja, from Test Board Data ................... 23
Thermal Characteristics for Different Package and PCB Configurations ............ 31
PowerPAD Package Template Description ........................................................ 35

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PowerPADTM Thermally Enhanced Package

PowerPAD Thermally Enhanced
Package

Abstract
The PowerPAO thermally enhanced package provides greater
design flexibility and increased thermal efficiency in a standard size
IC package. PowerPAO's improved performance permits higher
clock speeds, more compact systems and more aggressive design
criteria.
PowerPAO packages are available In several standard surface
mount configurations. They can be mounted using standard printed
circuit board (PCB) assembly techniques, and can be removed and
replaced using standard repair procedures.
To make optim um use of the thermal efficiencies designed Into the
PowerPAD package, the PCB must be designed with this
technology in mind. This document will focus on the specifics of
integrating a PowerPAO package Into the PCB design.

PowerPAD Thermally Enhanced Package

7

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5-45

PowerPADTM Thermally Enhanced Package

1.

Introduction
The PowerPAD concept is implemented in a standard epoxy-resin
package material. The integrated circuit die is attached to the
leadframe die pad using a thermally conductive epoxy. The package
Is molded so that the leadframe die pad Is exposed at a surface of
the package. This provides an extremely low thermal resistance (9,.)
path between the IC junction and the exterior of the case. Because
the external surface of the leadframe die pad is on the PCB side of
the package, it can be attached to the board using standard flow
soldering techniques. This allows eflicient attachment to the board,
and permits board structures to be used as heat sinks for the IC.
Using vias, the leadframe die pad can be attached to a ground plane
or special heat sink structure designed into the PCB. For the first
time, the PCB designer can implement power packaging without the
constraints of extra hardware, special assembly instructions, thermal
grease or additional heat sinks.

Figure 1. Schematic Representation of the PowerPAD Package Components
E (COPPER ALLOY)
IC (SILICON)
DIE ATTACH (EPOXY)

MOLD COMPOUND (EPOXY)
Section View of a PowerPAD(tm)
PACKAGE

Because the exact thermal performance of any PCB is dependent
on the details of the circuit design and component installation, exact
performance figures cannot be given here. However, representative
performance is very important in making design decisions. The data
shown in Table 1 is typical of the performance that can be expected
from the PowerPAD package.

8

SLMAOO2

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PowerPADTM Thermally Enhanced Package

Table 1. Typical Power Handling Capabilities of PowerPAD Packages
Standard Package

PowerPAD Package

0.75 W

3.25 W
2.32 W

0.55 W

Assumes 150° C junction temperature and 800 C ambient temperature.
Values are calculated from 9)& figures shown in Appendix A.

For example, the user can expect 3.25 watts of power handling
capability for the PowerPAD version of the 2Q.pin SSOP package.
The standard version of this package can only handle 0.75 watts.
Details for all package styles and sizes are given In Appendix A.

PowerPAD Thermally Enhanced Package

9

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5-47

PowerPADTM Therma"y Enhanced Package

2.

Installation and Use

2.1 PCB Attachment
Proper·thermal management of the PowerPAD package requires
PCB preparation. This preparation Is nat difficult, nor does it use any
extraordinary PCB design techniques, however it is necessary for
proper heat removal.

Figure 2. Bottom and Top View of the 20 pin TSSOP PowerPAD Package

20 PIN TSSOP, PowerPAD(tm)
PACKAGE
RELEASED FOR VOLUME
PRODUCTION SEPrEMBER,
1995.

10

SLMAOO2

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PowerPADTM Thermally Enhanced Package

Fl9ure 3. 64 Pin, 14 x 14 x 1.0mm Body TQFP PowerPAD Package

All of the thermally enhanced packages incorporate features that
provide a very low thermal resistance path for heat removal from the
integrated circuit - either to and through a printed circuit board (in
the case of zero airflow environments), or to an external heatsink.
The TI PowerPAD implementation does this by creating a leaclframe
where the bottom of the die pad is even with a surface of the
package (as opposed to the case where a heat slug is embedded in
the package body to create the thermal path). (See Figure 2 and
Figure 3.)

2.2 PCB Design Considerations
The printed circuit board that will be used with PowerPAD packages
must have features included in the design to remove the heat from
the package efficiently.

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

As a minimum, there must be an area of solder-tinned-copper
underneath the PowerPAD package. This area Is called the thermal
land. As detailed below, the thermal land will vary in size depending
on the PowerPAD package being used, the PCB construction and
the. amount of heat that needs to be removed. In addition, this
thermal land may or may not contain thermal vias depending on
PCB construction. The requirements for thermal lands and thermalvias are detailed below.

2.3 Thermal Lands
A thermal land is required on the surface of the PCB directly
underneath the body of the PowerPAD package. During normal
surface mount flow solder operations the leadframe on the
underside of the package will be soldered to this thermal land
creating a very efficient thermal path. Normally, the PCB thermal
land will have a number of thermal vias within it that provide a
thermal path to internal copper areas (or to the opposite side of the
PCB) that provide for more efficient heat removal. The size of the
thermal land should be as large as needed to dissipate the required
heat.
For simple, double-sided PCBs, where there are no internal layers,
the surface layers must be used to remove heat. Shown in Figure 4
is an example of a thermal land for a 24-pln package. Details of the
package, the thermal land and the required solder mask are shown.
If the PCB copper area Is not sufficient to remove the heat, the
dasigner can also consider external means of heat conduction, such
as attaching the copper planas to a convenient chassis member or
other hardware connection.
Figure 4. Package and PCB Land Configuration for a Single Layer PCB

24-Pin P'w'P TherMCll LClyout Single Lo.yer

0"
7.70

24-pln PIJP Po.cknge

Lnnd Po.ttern

BottOM ViE'1I'

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INSTRUMENTS
POST OFACE BOX 655303 • DALLAS. lCXAS 75265

Solder MQsl<

PowerPADTM Thermally Enhanced Package

For multilayer PCBs, the designer can take advantage of internal
copper layers (such as the ground plane) for heat removal. The
external thermal land on the surface layer is still required, however
the thermal vias can conduct heat out through the internal power or
ground plane. Shown in Figure 5 is an example of a thermal land
used for multilayer PCB construction. In this case, the primary
method of heat removal is down through the thermal vias to an
internal copper plane.

Figure 5. Package and PCB Land Configuration for a Multi-Layer PCB

24-Pin P,,",P TherMnl Lnyout Multi-Luyer

•"~~~=<~dJW""..r""'-

~'M

7,10l1li

"'"

0.191'11

IIk-~7tM-:J
:~:: II
6.lC

~~

BottDMlJiII1

""' ...

lmndPattl:'rn

2.f-pinPVPPo.clmge

SoIdEorMlSk
[onpSide

Shown in Figure 6 are the details of a 64 pin TOFP PowerPAD
package. The recommended PCB thermal land for this package is
shown in Figure 7.
The maximum land size for TOFP packages is the package body
size minus 2.0 mm. This land is normally attached to the PCB for
heat removal, but can be configured to take the heat to an external
heat sink. This is preferred when airflow is available.

PowerPAD Thermally Enhanced Package

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POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

5--51

PowerPADTM Thermally Enhanced Package

Figure 6. 64 pin TQFP Package wfth PowerPAD Implemented, Bottom View

64-PAP PowerPADCtM) PACKAGE

64-Pln PAP Pockage
Bottofl'l View

Figure 7. PCB Thermal Land Design Considerations for Thermally Enhanced
TQFP Packages
Multi-Lo.y<>r

Ll1ndPo.ttern
COMP $Ide

14

SDlder Mo.sk
CaMp Side

SLMAOO2

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INSTRUMENTS
POST OFFICE BOX 655303 • DALlAS. TEXAS 75265

PowerPADTM Thermally Enhanced Package

2.4 Thermal Vias
Thermal vias are the primary method of heat transfer from the PCB
thermal land to the internal copper planes or to other heat removal
sources. The number of vias used, the size of the vias and the
construction of the vias are all important factors in both the
PowerPAD package thermal performance and the package-to-PCB
assembly. Recommendations and guidelines for thermal vias follow.
Shown in Figure 8 and Figure 9 are the effects on PCB thermal
resistance of varying the number of thermal vias for various sizes of
die for 2- and 4-layer PCBs. As can be seen from the curves, there
Is a point of diminishing returns where additional vias will not
significantly Improve the thermal transfer through the board. For a
small die, having from five to nine vias should prove adequate for
most applications. For larger die, a higher number may be used
simply because there is more space available under the larger
package. Shown in Figure 10 are examples of ideal thermal land
size and thermal via patterns for PowerPADTM packages using
O.33mm (13 mil) diameter vias plated with 1 oz. copper. This thermal
via pattern set represents a copper cross section in the barrel of the
thermal via of approximately 1% of the total thermal land area.
Fewer vias may be utilized and still attain a reasonable thermal
transfer into and through the PCB as shown in Figures 8 and 9.
The number of thermal vias will vary with each product being
assembled to the PCB, depending on the amount of heat that must
be moved eway from the package, and the efficiency of the system
heat removal method. Characterization of the heat removal
efficiency versus the thermal via copper surface area should be
performed to arrive at an optimum value for a given board
construction. Then the number of vias required can be determ ined
for any new design to achieve the desired thermal removal value.

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

Figure 8. Impact of the Number of Thermal Vias versus Chip Area (Die Area)
JEDEC 2·LAYER BOARD THERMAL RESISTANCE (JC)
COMPARISON

VIAS

1

2

3

4

5

6

7

8

9

10

THERMAL VIAS COPPER CROSS AREA (% OF DIE AREA)
Note: Apply bare die 10 the JEDEC board

Figure 9. Impact of the Number of 0.33mm (0.013 inch) Diameter Thermal Vias
versus Chip Area (Die Area)
JEDEC 4·LAYER BOARD THERMAL RESISTANCE (JC)
vs THERMAL VIAS CROSS AREA

THERMAL VIAS COPPER CROSS AREA (% OF DIE AREA)
Note: Apply bare die to the JEDEC board

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POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

PowerPADTM Thermally Enhanced Package

.I'

Figure 10. Ideal Thermal Land Size and Thermal Via Pattems for PowerPAD

'm'.:..
.L
I:·

12.6 • •

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TSSOP

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axB

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ax.

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axil

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fl.tl

~sop
6 IliI apwp

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11

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rii1

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~
liijJ TSSOP

8.43

U.7~

~WP

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...':1'
~

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ex.

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ex.
....j-.j-M

00.

-I-M-IThermal vias connect the thermal land to internal or external copper
planes and should have a drill diameter sufficiently small so that the
via hole is effectively plugged when the barrel of the via is plated
with copper. This plug is needed to prevent wlcking the solder away
from the Interface between the package body and the thermal land
on the surface of the board during solder reflow. The experiments
conducted jointly with Solectron Texas indicate that a via drill
diameter of 0.33mm (13 mils) or smaller works well when 1 ounce
copper Is plated at the surface of the board and simultaneously
plating the barrel of the via. If the thermal vias will not be plugged
when the copper plating is performed, then a solder mask material
should be used to cap the vias with a dimension equal to the via
diameter + 0.1 mm minimum. This will prevent the solder from being
wicked through the thermal via and potentially creating a solder void
In the region between the package bottom and the thermal land on
the surface of the PCB.

PowerPAD Thermally Enhanced Package

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5-55

PowerPADTM Thermally Enhanced Package

To assure the optimum thermal transfer through the thermal vias to
internal planes or the reverse side of the PCB, the therm aI vias used
in the thermal land should not use web construction techniques.
Web construction on PCB vias is a standard technique used in most
PCBs today to facilitate soldering, by constructing the via so that it
has a high thermal resistance. This Is not desirable for heat removal
from the PowerPAD package. Therefore it Is recommended that all
vias used under the package make Internal connections to the
planes using a continuous connection completely around the hole
diameter. Web construction for thermal vias is not recommended.

2.5 Solder Stencil Determination
A series of experiments were conducted at Solectron-Texas to
datermlne the effects of solder stencil thickness on the quality of the
solder joint between the thermal pad of a PowerPAD package and
the thermal land on the surface of the PCB. Stencil thickness of 5, 6,
and 7 mils were used in conjunction with a metal squeegee to
deposit solder In the desired locations on the board. Note: 6 and 7
mil thick solder stencil Is normally used with package lead pitch of
0.5 and O.65mm respectively. A 5 mil thick stencil is normally used
for packages with O.4mm lead pitch to avoid solder bridging during
reflow.
It was found that the standoff height for the package being attached
to the PCB was critical In making good solder Joints between the
thermal pad of the package and the thermal land on the PCB. Note:
during this series of experiments, a good solder joint was defined as
a connection that joined at least 90% of the area of the smallest
pattern to its intended connection point - such as the thermal pad of
the package to the thermal land on the PCB. When the standoff
height of the package (i.e., the distance between the bottom of the
package leads and the bottom of the package body) was in the
range of 0 to 2 mils, the paCkage tended to float on the solder. This
led to the possibility that all leads of the package would not be
soldered to the lead traces on the board. This happened even when
the 5 mil thick stencil was utilized. There were also cases when the
solder was squeezed out from the desired land area, and then
formed solder balls during the reflow process - an undesirable result
that could cause shorting between package leads on the board
surface, or short the thermal land on the PCB to the lead traces. A
standoff height of 2.0 to 4.2 mils provided good solder joints for both
the leads and the thermal pad for stenCil thickness of 5, 6, and 7
mils. When the standoff height of the package was between 4.2 and
6.0 mils, only the 6 and 7 mil thick stencil provided consistently good
solder joints for both the package leads and the thermal-pad to
thermal-land bond. A general guideline would be to use the thickest
solder stencil that works well for the products being assembled for
the most process margin in assembling thermally enhanced parts to
a PCB.

18

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PowerPADTM Thermally Enhanced Package

The Joint Electron Devices Engineering Council (JEDEC)
specification for the standoff height of TSSOP and TQFP packages
is the range of 0.05 to 0.15mm (1.97 to 5.91 mils), and is an'
acceptable range when the solder stencil thickness of 6 and 7 mils
are used. Texas Instruments has elected to center the stand-off
height of the Power PAD packages at 3.5 mils (within the JEDEC
specification range) to provide good package to PCB solder joint
characteristics for standard solder stencil thickness of 5, 6, and 7
mils - the most common range within industry practice tOday.

PowerPAD Thermally Enhanced Package

19

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INSTRUMENTS
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PowerPADTM Thermally Enhanced Package

3.

Assembly
Solder joint inspection in the attachment area of the thermal pad of
the thermally enhanced packages to the thermal land on the PCB is
difficult to perform with the best option to date being x-ray
Inspection. Tests performed within Texas Instruments and during the
joint PCB experiments with Solectron-Texas indicate that x-ray
Inspection will allow detection of voiding within the solder joint and
could be used either in a monitor mode, or for 100% inspection if
required by the application. However, this is a slow and costly
process so an effort was made to determine the minimum amount of
solder required in this joint before degradation of the thermal
performance became significant.
The experimental vehicle used in determining the amount of solder
required was a 6S2P double sided test board with copper thermal
lands on the surface of the board representing 0%, 7.5%, 22%, and
83% of the package body area. The package used was a 100 pin
PowerPAD package (side B - standard enhanced Vf side of the
PCB) as shown in Figure 11. There was additional copper area on
the surface of the A side of the board due to connections between
selected pins and the thermal land area. Four thermal vias were
created In each therm aI land area with connections to the Internal
power or ground plane, and continuing to make connection to the
thermal land on the opposite side of the board.
A thermal test chip (Texas Instruments X-1158240) with dimensions
of 6.1 mm (O.240-lnch) square was assembled in the test packages
using die pad sizes of 6.0mm square, and 9.0mm square. The
assembled unitS were then mounted to the PCB using either eutectic
Sn63:Pb37 solder or thermally conductive epoxy adhesive.
Measurement of the thermal resistance junction-ta-case and thermal
resistance junctlon-to-ambient with the individual packed parts
powered at 2.5 watts was made using standard techniques for these
measurements. Results are shown in Table 1 for tests with and
without attachment between the package thermal pad and the board
thermal land, as well as a comparison between solder and thermally
conductive epoxy attachment. Table 2 provides the effective
connection area obtained for each of the measurement points.

20

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INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

PowerPADTM Thermally Enhanced Package

Figure 11. Test Board for Measurement of B}c and Bla Using 100 pin PowerPAD
TQFP Packages

THERMAL TEST BOARD LAYOUT
2 SIDED, 8 LAYER BOARD
=m.~ EI=:r~~~~~1

=~l:.

=lm.~

=~~,.

1IIItTt. . .
IEIlSTOllM

=m&t"L~t--'--""'"

EIIIITE.II

IIE$ISTDIH

=m:. fa
4

LAYERS 1,2,3,6,7,8 ARE 1 OZ COPPER, 20% COVERAGE
LAYERS 4, 5 ARE 1 OZ COPPER, 80% COVERAGE
VIAS IN BOARD CONNECT COMMONS FROM TOP TO LAYERS 4 AND 5
ANTICIPATED POWER LEVEL OF 2.5 WATT MAX FOR EACH PART
STANDARD THERMAL TEST BOARD DIMENSIONS
CONNECTOR IS 0.125 INCH PITCH, 18 CONTACTS/SIDE, 2 SIDES
PACKAGE IS LQFPrrQFP 14 X 14 X 1.0 OR 1.4mm BODY SIZE; 0.5mm LEAD PITCH
VENDOR .. SERIUS SOLUTIONS (RAY MULl1NS 404-9748) NUMBER 10-00001-008 LAYER; K FACTOR X 8; 100 LQFP{TQFP

The relative thermal land size and location is shown along with the
location of the therm al vias that connect the surface thermal land to
the internal power or ground plane, and continuing to connect to the
thermal land on the opposite side of the board. The board is
approximately B2.5mm (3.25 inch) square.
Table 2 and Table 3 show the thermal resistance data for Sjc and Sja
Qunction to case, and junction to ambient) for the B layer thermal
test board, with the copper thermal land on the PCB shown as a
percentage of the area of the package body.

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

Table 2. Measured 8jc from Test Board
MEASURED DATA
Part position

on PCB

PCB Copper land

90

~o

6mm Die Pad

9mm Die Pad

~
9mm Die Pad

9mm DIe Pad

~o
9mm Die Pad

9'0

as % of package

Soldered one

Not Soldered

Soldered one

Soldered both

Epoxy used 10

body area

side only

to PCB

side only

s~ofPCB

_ 1 0 PCB

0
7.5
22
83
0
7.5
30
85

9.3

9.9
7.2
8.3
6.2
9.1
6.3

11.4
5.8
7.2
6.2
7.8
6.B
6.6
6.4

7.2
7.5
6.2
7.B
6.B
6.5
6.9

16
46
26
36
2A

3A
1A
4A

8.B
6.2
8.7
7.6

7.4
8.3
8
7.3

7.5

Notes. 1)
.2)
3)

6.4

Numbers In bold have die pad attached to the board .
Power level for all measurements is 2.5 watt.
9)0 is measured in 1 cubic foot of liquid freon.

Table 3. Measured 8ja from Test Board
MEASURED DATA

9,

9,

9,

~,

Part position

PCB Copper land

6mm Die Pad

9mm Die Pad

9mmDiePad

9mm Die Pad

9mm DiaPed

on PCB

as % of peckage
body area

Soldered one
side only

Not Soldered

Solderedone
side only

Soldered both
sides of PCB

Epoxy used 10
attach 10 PCB

0
7.5
22
83
0
7.5
30
85

33.B

40.6
27
25.B
26.9
33.3
24.4

44.3
23.1
25
24.6
32.3
24.9
24.4
24.6

25.5
24.3
24
25.B
25.2
23.2
24

16
46
26
36
2A

3A
1A
4A

9.

2B.4
24.2
34.4
33.5
33.3

Notes: 1)
2)

3)

10 PCB

34
33
31
30

25.5

Numbers In bold have die pad attached to the board.
Power level for ali measurements is 2.5 Walt.
8ja is measured in 1 cubic foot of still air.

Small changes in the percentage of copper land area (between the
"A" side of the PCB and the "B" side of the PCB) do not significantly
affect the therm al resistance.

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PowerPADTM Thermally Enhanced Package

Table 4 and Table 5 show the relationship of the solder joint area
between the thermal pad in the PowerPAD package and the thermal
land of the PCB for the thermal resistance values obtained in Table
2 and Table 3.

Table 4. Relationship of the Solder Joint Area on €Jle, from Test Board Data
THERMAL PAD TO THERMAL LAND CONNECTION AREA ANALYSIS - %
~.

~.

~

9.

~.

Position

PCB Copper land

6mm Ole Pad

9mm Die Pad 9mm Die Pad 9mmOiePad

on PCB

slze on PCB

Soldered one

Not Soldered Soldered one Soldered both Epoxy used to

16
46
26
36
2A
3A
lA
4A

0
4*(2)<2)
1*(6x6)
1*(12x12)
0
4*(2)<2)
1*(6x6)+4*(5.7)
1*(12x12)+4*(5.6)
Notes: 1)
2)
3)

9mm Die Pad

side only

to PCB

side only

.~ofPCB

altach to PCB

0
36
80
100
0
80
85
100

0
16
32
100
0
16
58
100

0
16
32
100
0
16
58
100

0
16
32
100
0
16
58
100

0
100
100
100
0
100
100
100

Numbers In bold have die pad attached to the board.
Power level for all measurements is 2.5 waH.

s" Is measured In 1 cubic foot of liquid fmon.

Table 5. Relationship of the Solder Joint Area on €Jja, from Test Board Data
THERMAL PAD TO THERMAL LAND CONNECTION AREA ANALYSIS - %

9.

9.

9 ..

a.._

~.

Position

PCB COpper land

6mm Die Pad

9mm Die Pad 9mm Die Pad 9mm Die Pad

9mm Die Pad

onpeB

as % of package

Soldered one

Not Soldered

Soldered one Soldered both

Epoxy used to

body area

side onty

to PCB

side only

sides of PCB

altach to PCB

a

0
36
80
100

a

0
16

0
16
32
100
0
16
58
100

0
100
100
100
0
100
100
100

16
46
26
36
2A
3A
lA
4A

4*(2)<2)
1*(6x6)
1*(I2xI2)

16
32
100

a

a

a

4*(2)<2)
1*(6x6)+4*(5.7)
1*(12x12)+4*(5.6)

80
85
100

16
58
100

Notes: 1)
2)
3)

32

100
0
16
58
100

Numbers In bold have die pad attached to the board.
Power level for all measurements is 2.5 waH.
9/a is measured in 1 cubic foot of stili air.

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

In this example, there is significant improwment in thermal heat
removal with solder joint areas as small as 16%, and the thermal
removal efficiency as measured by 9Jc and 9ja are within
measurement error tolerance for all solder joint areas greater than
32%.
Based on the measured data for this test board configuration, Texas
Instruments recommends a minimum solder joint area of 50% of the
package thermal pad area when the part Is assembled on a PCB.
The results of the PCB assembly study conducted with SolectronTexas indicate that standard board assembly processes and
materials will normally achiew >80% solder joint area without any
attempt to optimize the process for thermally enhanced packages. A
characterization of the solder joint achlewd with a glwn process
should be conducted to assure that the results obtained during
testing apply directly to the customer application, and that the
thermal efficiency in the customer application is similar to the
thermal test board results for the power lewl of the packaged
component. If the heat removal is not at the efficiency desired, then
either additional thermal via structures will haw to be added to the
PCB construction, or additional thermal removal paths will need to
be defined (such as direct contact with the system chassis).
An altematlw to attaching the therm aI pad of the package to the
thermal land of the PCB with solder is to use thermally conductiw
epoxy for the attachment. This epoxy can either be dispensed from
the liquid form with a material that will cure during the refiow cycle,
or a 'B' staged preform that will raceiw the final cure during the
reflow cycle. These materials can be the same as normally used
with extemally applied heat sinks. When epoxy is used as the
attachment mechanism, then the effectiw attachment area Is 100%
of the die pad area, and there is some added benefit as thermal
transfer to the PCB can occur, ewn with no copper thermal land at
the surface of the PCB.

3.1 Solder Reflow Profile Suggestion
The refiow profile for IR board assembly using the Texas
Instruments PowerPAD packages does not haw to change from
that used with conwntional plastic packaged parts. The construction
of the package does not add thermal mass, and the only new
thermal load is due to the increased solder area between the
package thermal pad and the thermal land on the PCB. A typlcallR
own profile for fine pitch surface mount packages is shown in
Figure 11. for eutectic Sn63:Pb37 solder. Nitrogen purged,
conwction IR refiow will be advantageous for this part to PCB
assembly to minimize the possibility of solder ball formation under
the package body.

24

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POST OFFICE BOX 655303 • DAlLAS, TEXAS 75265

PowerPADTM Thermally Enhanced Package

Figure 12 shows a typical infrared (lR) oven profile for a fine pitch
plastic package assembly mounted to an FR-4 PCB using eutectic
Sn63:Pb37 solder.

Figure 12. Typical Infrared Oven Profile
300~----rl-rI~I~--~I~I--~I--~I~I~--~I~I---rI---'IIr---~I--~I~Ir------'

270

I I I I II I I I II I II
I
II
t---~I-t-i----H---t--i--t---tt--t---H---~-"------

240
210 I
I I
II
I
I I
I I
I
II
~ ISO -I-----l--I--+---H---~--++---H---+-~ 150 I
I I
I
I
I I
I I
I
II
Il
I I I
II
I
I
II
I
II
I
I I
Fi 120
-T-t---tt--t---"1j---~-1t----90 I
I
II
I
I I
I I
I
II
I
I I
I
I
II
I
I I
II
I
II
I
I I

II

II

I

t--1-t-T---H--- ,
~

t-~-i---~---~--~--r---~~--t---1---1r-ii-----I I
12 I
0.6

I
13

II
I

I
15

I
I

I
I

II
I

1.2
1.9
Max Slope: ·3.2

I
IS

II
19

I
II
110 I I
3.1
3.7
Seconds over 183: 4S

4.4
Time

Belt Speed = 3S.00 inches/minute
ZONE SET POINTS
1
160

2
125

9
265

10
260

3
115

4
110

5
190

7
160

7
160

8
190

Peak temperature should be approximately 220 degrees centigrade,
and the exposure time should normally be less than 1 minute at
temperatures above 183 degrees centigrade.

3.2 Installation and Assembly Summary
The PowerPAD package families can be attached to printed circuit
boards using conventional Infrared solder reflow techniques that are
standard in the industry today without changing the refiow process
used for normal fine pitch surface mount package assembly. A
minimum solder attachment area of 50% of the package thermal pad
area is recom m ended to provide efficient heat rem oval from the
semiconductor package, with the heat being carried into or through
the PCB to the final thermal management system. This attachment
can be achieved either by the use of solder for the joining material,
or through the use of thermally conductive epoxy materials. Typical
PCB thermal land pattern definitions have been provided that have
been shown to work with 4 and 8 layer PCB test boards, and can be
extended for use by other board structures.

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

.
4.

Repair
Reworking thermally enhanced packaged semiconductors that have
been attached to PCB assemblies through the use of solder or
epoxy attachment can present significant challenges, depanding on
the point at which the re-work is to be accomplished. Tests of rework procadures to date Indicate that part removal from the PCB Is
succassful with all of the conventional techniques used in the
industry today. The challenge is part replacement on the board due
to the combined thermal enhancement of the PCB itself, and the
addition of thermal removal enhancement features to the
semiconductor package. The traditional steps in the rework or repair
process can be simply identified by the following steps for solder
attached components:

1) Unsolder old component from the board
2) Remove any remaining solder from the part location
3) Clean the PCB assembly

4) Tin the lands on the PCB and leads, or apply solder paste to the
lands on the PCB
5) Target, align, and place new component on the PCB
6) Reflow the new component on the PCB

7) Clean the PCB assembly
When thermally conductive epoxy has been used to attach the
thermal pad of the package to the thermal land on the PCB, the
same basic steps In the rework or repair procedure can be followed
with only minor modifications:

1) Unsolder old component and torque package to remove from the
board
2) Remove any remaining solder from the part location
3) Remove any remaining epoxy from the thermal land on the PCB

4) Clean the PCB assembly
5) Tin the lands on the PCB and leads, or apply solder paste to the
lands on the PCB
6) Place new thermally conductive "B" staged epoxy preform or

dispense epoxy on thermal land

7) Target, align, and place new component on the PCB
8) Reflow the new component on the PCB

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PowerPADTM Thermally Enhanced Package

9) Complete epoxy cure (if required as a separate step)
10) Clean the PCB assembly

4.1 Part Removal From PCBs
Almost any removal process will work to remove the device from the
PCB, even with the thermal pad of the package soldered to the
PCB. Heat is easily transferred to the area of the solder attachment
either from the exposed surface thermal lend of the PCB (single
layer example), or through the thermal vias in the PCB (multi-layer
example) from the backSide of the PCB.
Re-work has been performed for both the TSSOP and TOFP
PowerPAD style packages using METCAL removal irons and hot air.
The specific example of a 20 pin TSSOP PowerPAD part removal is
discussed in detail.
A 750-Watt METCAL removal iron was used In conjunction with hot
air to verify the removal method efficiency to take 20 pin PowerPAD
TSSOP packages off of assembly test boards. The hot air method Is
recommended as it subjects the PCB and surrounding components
to less thermal and mechanical stress than other methods available,
and has been proven to be much easier to control than. some of the
hot bar techniques. Use of the hot air method may require
assemblers to acquire tools specifically for the smaller packages
since most assemblers use a hot bar method for packages of this
size. (Note: This same tool will also be needed for part reattachment to the PCB when the hot air method is employed). A tool
with an integrated vacuum pick up tip will be an advantage in the
part removal process so the part can be physically removed from the
board as soon as the solder reaches liquidus. Preheating of the local
area of the PCB to a temperature of approximately 160 degrees
centigrade can make the part removal easier. This is especially
helpful in the case of larger packages such as 56 pin TSSOP or
1OQ-pin TOFP style packages. This preheat will be required in the
thermal removal method if the semiconductor package is a heat slug
package rather than the TI PowerPAD package version. Some
experimentation will be required to find the optimum procedure to
use for any specific PCB construction and thermally enhanced
package version.
After the part has been removed from the PCB, conventional
techniques to clean the area of the part attachment - such as solder
wicking - will be needed to prepare the location for subsequent
attachment of a new component.

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

When thermally conductive epoxy has been used for attachment of
the package thermal pad to the thermal land on the PCB, a slightly
different approach to part removal must be used, This will require a
tool that has dimensions that will allow contact with the sides of the
package body directly above the leads, and will allow the package to
be twisted or rotated horizontally when the solder joints of the
package leads have reached liquidus. The temperature at the epoxy
interface to the package thermal pad or the PCB thermal land must
be above the glass transition tem perature of the epoxy (typically less
than 180 degrees centigrade) to break the adhesion between the
epoxy and the attach location with the twisting or rotational method
discussed above. In most cases, any remaining epoxy on the PCB
after part removal can be removed by peeling it from the surfaceoccasionally, it will be necessary to apply heat to the epoxy location
so it will peel away from the PCB cleanly.

4.2 Attachment of a Replacement Component to the PCB
Preparation of the PCB for attachment of a new component follows
normal industry practice with respect to the lands on the board and
the leads of the package. Both may be tinned, and/or solder paste
applied to the lands for new component attachment. In addition,
when solder will be used to re-attach the thermal pad of the package
to the thermal land on the PCB, solder paste will need to be applied
to the surface of the thermal land on the board. This may be in the
form of stripes of solder paste with sufficient volume to achieve the
desired solder coverage, or a solder preform may be applied to the
location for attachment. In a factory environment, the component is
then placed in the desired location and alignment, and processed
through a reflow oven to re-establish the desired solder joints. This
is the most desirable process and is normally the easiest to
accomplish.
When a manual or off-line attachment and reflow procedure is to be
used, the challenge of supplying sufficient heat to the components
and solder becomes a greater concern. In most cases, the corner
leads of the package being attached will be tack soldered to hold the
component in alignment so the balance of the leads and the thermal
pad to therm al land solder reflow can be accom plished without
causing part movement from its desired location. As in the part
removal case, it is advisable to pre-heat the board or the specific
device location to a temperature below the melting point of the
solder to minimize the amount of heat that must be provided by the
reflow device as the part is being attached. A good starting point is
to pre-heat to approximately 160 degrees centigrade. A hot gas
reflow tool can then be used to complete the solder joint formation
both at the leads and for the connection of the therm al pad to the
thermal land of the PCB. Care must be taken at this operation to
avoid blowing solder out from the thermal pad to thermal land
interface and causing solder balling under the package or creating

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PowerPADTM Thermally Enhanced Package

lead to lead or thermal land to lead shorts. The thermal
enhancement of the package and the PCB will require a higher
temperature gas or higher gas flow to reach solder liquidus than
would be needed with an assembly lacking these enhancements.
The 1001 should be specifically sized 10 Ihe part being reworked to
minimize possible damage to surrounding components or Ihe PCB
ilself.
If the re-attachment of the interface between the thermal pad of the
package and the thermal land of the PCB using solder attachment is
too difficult to control using hot gas methods, then the best approach
is to use either a thermally conductive "B' staged epoxy preform cut
to the shape of the thermal land on the PCB, or dispensing liquid
thermally conductive epoxy in a pattern on the thermal land thai will
result in at least a 50% void free connection between the pad and
Ihe land. Virtually any epoxy material that is used for Ihe attachment
of external heat sinks 10 packaged com ponents is suitable for this
application, and cure lime/temperature requirements can be
matched to the product need (anywhere from 24 hours at room
tem perature to less Ihan 1 hour at lem peratures below 100 degrees
centigrade). Care must be taken to choose a material with limited
run-out to avoid the possibility of shorting adjacent package leads
together or shorting the thermal land of the PCB to the package
leads.
It should be noted that the Texas Instruments PowerPAD packages
are easier to rework at the board level than other semiconductor
packages utilizing melal slugs for the thermal path between the chip
and the PCB. This is due to the additional requirement for heating
the total mass of the slug to reflow tem peratures versus heating the
thermal pad of the PowerPAD package. The hot gas temperature
and/or flow becomes critical for effective joining of the components
without causing damage to the adjacent components or the PCB. In
either case, the use of thermally conductive epoxy materials will
make the rework task easier and more reliable to perform In a
manual repair environment.

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

5.

Summary
An overview of the design, use and performance of the Texas
Instruments PowerPAD packege has been presented. The packege
Is Simple to use and can be assembled and repaired using existing
assembly and manufacturing tools and techniques. Package
performance is outstanding. By exposing the leadfreme on the
peckege bottom, extremely effiCient thermal transfer between the die
and the PCB cen be achieved.
The simplicity of the PowerPAD peckage not only makes for a low
cost package, but there is no additional cost In labor or material for
the customer using standard surface mount assembly techniques.
The only preperation needed to implement a PowerPAD design Is at
the PCB design stage. Simply by including a thermal land and
thermal vias on the PCB the design can use the PowerPAD package
effectively.

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PowerPADTM Thermally Enhanced Package

Appendix A. Thermal Modeling of Power PAD Packages
Table 6. Thermal Characteristics for Different Package and PCB Configurations
2 oz. Trace and COpper Pad 2 oz. Trace and COpper Pad
wHhout Solder
wlthSOldor

Package Descrlp1lon

Pkg
Type

I

I

Package
DHignalOr

0...
rclW)

20
24
28

DWP
DWP

21.46
20.77

DWP

TVSOP

80
100
20
24
48
56

TSSOP

48
56
64
28
30
32
38
28
30
38
44
50
14
16
20
24
28

SSOP

TQFP

LQFP

Pin

I

I

I('CiW)

92.95
80,49
69.73

16.58
13,49
11.24

2.212
1,959
1.641

0,359
0,313
3,318
3,176
1,138
0,99

65,53
54,55
192,65
179.91
107,49
95,48

4,69
3,73
28,85
28.41
12,32
10.40

0,353
0,297
1,054
0,999
0,58
0,526

0,443
0.401
0,357
0,556
0,551
0.468
0,444
1,424
1.408
1,13
0.962
0.892
2.711
2.6

6.63
5.81
4,69
8,96
8,73
7,32
6,57
16,13
16,05
12.42
10.47
9.34
26.88
26.56
19.90
14.63
12.41

0,434
0,395
0,35
0,548
0._

32,64
28,45
95,88
89,50
52,82
46,69

0,21
0,17
2,46
2.46
0,72
0,58

0.22
0,212
0,196
0,244
0,233
0,233
0,219
0,534
0,532
0,447

40,27
36.42
32,52
51,28
48,34
44,32
41,18
63,99
63,32
52,93

0,32
0,27
0,21
0,45
0,45
0,32

0.406
0.369
0.851
0.848
0.607
0.489
0.446

47.18
43.76
97.65
90.26
74.41
62.05
56.21

1.14
0.38
0.12
0.38
0.17
0.12
0.12

0.429
0.192
0.155
0.19
0,174

0.13
0.10
0.10
0.10

19.52

43.91
38.43
33.92

DDP
DDP
DGP
DGP
DGP
DGP

19,88
18,35
37,92
38,87
27,35
25,42

0,21
0,17
2,46
2,46
0,72
0,58

0,196
0,182
1,074
1.056
0,45
0,406

DCA
DCA
DCA
DAP
DAP
DAP
DAP
DCP
DCP
DCP
DCP
DCP

22,30
21.17
19,89
25,10
24,20
23,51
22.41
30,62
30,55
27,41
25,57
24,10
37.47
36.51
32,63

PWP

30.13
27.87

0,32
0,27
0,21
0,45
0,45
0.32
0,31
0,94
0,94
0,72
0,58
0.51
2.07
2.07
1.40
0.92
0.72

48
52
64
64
80
100
128

PHP
PGP
PBP
PAP
PFP
PZP
PNP

29.11
21.61
17.46
21.47
19.04
17.28
17.17

144
178
180
208

PAP

15.68
14.52
11.14
10.96

PTP
PSP
pyp

0...
('CIW)

6.031
4,88
4.109

1.617
1.507
1.337

PWP
PWP

oz. trace

0.37
0.27
0.22

0.37
0.27
0,22

PWP

I

0...
rclW)

0...
(,CIW)

PWP

I

'i'"
(,CIW)

'i'"
('CIW)

Count

Standard Package

JEDEC Low EIhIct with 1

0...
rCIW)

0...
rclW)

I

'i'"

0.92
0.72

1.263
1.169

84,04
75,50
65,70
110,80
103,45
95,63
87.32
133,67
131,23
109,55
97.13
89,53
195.35
182.31
151.89
128.44
115.82

0.154
0.152

64.42
42.58
28.04
42.20
31.52
27.32
27.07

1,14
0.38
0.12
0.38
0.17
0.12
0.12

1.282
0.391
0.252
0.388
0.29
0,247
0.244

108,71
77.15
52.21
75.63
57.75
49.17
48.39

18.18
7,83
3.12
7.80
4.20
3.11
3.11

0.511
0.353
0.267
0.347
0.297
0.252
0.248

0.199
0,17
0.14
0.139

27.52
24.46
22.40
21.48

0.13
0.10
0.10
0.10

0.346
0.28
0.266
0,258

47.34
42.95
43.93
39.18

4.62
3.67
3.70
3.66

0.288
0.257
0.262
0.235

0.31

0,94
0,94
0,72
0.58
0.51
2.07
2.07
1,40

1.m

PowerPAD Thermally Enhanced Package

0.478
0,454
0,707
0.695
0,598
0.538
0.5
1.047
0.984
0,77
0.665
0.623

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General
Thermal modeling is used to estimate the performance and
capability of Ie packages. From a thermal model, design changes
can be made and thermally tested before any time is spent on
manufacturing. It can also be determined what components have the
most influence on the heat dissipation of a package. Models can
give an approximation of the performance of a package under many
different conditions. In this case, a thermal analysis was performed
In order to approximate the improved performance of a PowerPAO
thermally enhanced package to that of a standard package.

Modeling Considerations
There are only a few differences between the thermal models of the
standard packages and models for PowerPAD. The geometry of
both packages was essentially the same, except for the location of
the lead frame bond pad. The pad for the thermally enhanced
PowerPAD package is deep downset, so its locetlon is further away
from the lead fingers than a standard package lead frame pad. Both
models used the maximum pad and die size possible for the
package, as well as using a lead frame that had a gap of one lead
frame thickness between the pad and the lead fingers. The lead
frame thickness was:
TQFP/LQFP: 0.127 mm, or 5 milS
TSSOP/lVSOP/SSOP: 0.147 mm, or 5.8 mils
In addition, the board design for the standard package is different
than the PowerPAO. One of the most influential components on the
performance of a package is board design. In order to take
adventage of PowerPAO's heat dissipating abilities, a board must be
used that acts similarly to a heat sink and allows for the use of the
exposed (and solderable) deep downset pad. This is Texas
Instruments' recommended board for PowerPAO (see

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PowerPADTM Thermally Enhanced Package

Figure 13). A summary of the board geometry Is included below.

Texas Instruments Recommended Board for PowerPAD
0.062' thick
3' x 3' (for packages <27 mm long)
4' x 4" (for packages >27 mm long)
2 oz. copper traces located on the top of the board (0.071 mm
thick)
Copper areas located on the top and bottom of the PCB for
soldering
Power and ground planes, 1 oz. copper (0.036 mm thick)
Thermal vias, 0.33 mm diameter, 1.5 mm pitch
Thermal isolation of power plane

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

Figure 13. Texas Instruments Recommended Board (Side View)
Solder Pad

ponent Traces

..==--_______-.1.5038

·1.5748 mm ComponentTrace

_ _ _ _ _ _ _ _ _ _-41.0142 ·1.0502 mm Ground Plane
1.5748 mm

. .----------110.5:246.0.5606 mm Power Plane
_ _ _ _ _ _ _ _ _ _---"u.u·0.071 mm Board Base & Bottom Pad

Solder Pad (bottom trace)

The standard packages were placed on a board that is commonly
used in the industry today, following the JEDEC standard. It does
not contain any of the thermal features that are found on the Texas
Instruments recommended board. It only has component traces on
the top of the board. A summary of the standard is located below:

JEDEC Low Effective Thermal Conductivity Board (Low-K)
0.062" thick
3" x 3" (for packages <27 mm long)
4" x 4" (for packages >27 mm long)
1 oz. copper traces located on the top of the board (0.036 mm
thick)
These boards were used to estimate the thermal resistance for both
PowerPAD and the standard packages under many different
conditions. While the PowerPAD can be used on a JEDEC low-k
board, in order to achieve the maximum thermal capability of the
package, it is recommended that it be used on the Texas
Instrumynls heat dissipating board design. It allows for the exposed
pad to be directly soldered to the board, which creates an extremely
low thermal resistance path for the heat to escape.

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PowerPADTM Thermally Enhanced Package

A general modeling template was used for each Power PAD
package, with variables dependent on the package size and type.
The package dimensions and an example of the template used to
model the packages are shown in Figure 14 and Table 7. While only
1/4 of the package was modeled (in order to simplify the model and
to lessen the calculation time), the dimensions shown are those for a
full model.

Figure 14. Thermal Pad and Lead Attachment to a PCB Using the PowerPAD
Package
Mold
compound

Table 7. PowerPAD Package Template Description
(A) PCB Thickness:
PCB Length:
PCB Width:
(B) Chip Thickness:
Chip Length:
Chip Width:
(C) Die Attach Thickness:
(D) Lead Frame Downset:
Tie Strap Width:
(E) PCB to Package Bottom:
(G) Shoulder Lead Width:
(H) Shoulder Lead Space:
(J) Shoulder to PCB DiS!.:
Not..: 1)
2)
3)

(I<) Package Thickness:
Package Length:
Package Width:
(L) Pad Thickness:
Pad Length:
Pad Width:
PCB Trace Length:
PCB Trace Thkn:
PCB Backplane Th:
PCB Trace Width:
(M) Foot Width:
(N) Foot Length on PCB:

1.S748mm
76.2mm (1)
76.2mm (1)
O.267mm
(2)
(2)
O.0127mm
(3)
(3)
0.09 mm
(3),(5),(6)
(3),(6)
(7)

(3)
(3)
(3)
O.147mm (8)
(3)

(3)
2S.4mm
0.071 mm
O.Omm (4)
0.254mm
(5)
(3)

99.6mm for packages > 27mm max length

Chip size Is 10 mils smaller than the largest pad size (5 mils from each side)
Dependent on package size and type
4) The recommended board requires the addition of two internal copper planes, solder pads, and
thermal vfas
5) Foot width was set equal to shoulder lead width for model efficiency
6) Lead p~ch Is equal to the shoulder lead width piUS the shoulder lead space (pitch = G + H)

PowerPAD Thermally Enhanced Package

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PowerPADTM Thermally Enhanced Package

7)

The shoulder to board distance Is equal to the downsat plus the board to package bottom distance (J
=D+E)

81 The pad thickness for TQFPILQFP Is equsl to 0.127 mm
9) All dimensions ara in mlUlmetelll.

In addition to following a template for the dimensions of the
package, a simplified lead frame was used. A description of the lead
frame geometry is seen In Figure 15.

Figure 15. General Leadframe Drawing Configuration

NOTE:

=

The lead frame downset bend area 20 mils (lead frame
thickness). For SSOP, TSSOP, and TVSOP packages,
add the bend area to the width of the pad. For TOFP and
LOFP, add the bend area to both the width and length of
the pad.

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PowerPADTM Thermally Enhanced Package

Results
The purpose of the thermal modeling analysis was to estimate the
increase In performance that could be achieved by using the
PowerPAD package over a standard package. For this package
comparison, several conditions were examined:
Case 1. PowerPAD soldered to the TI recommended board
Case 2. PowerPAD not soldered to the TI recommended board
Case 3. A standard package configuration on a low-k board
Case 4. A standard package on the TI recommended board
The first three cases show a comparison of PowerPAD packages on
the recommended board to standard packages on a board
commonly used In the Industry. The results are shown in Table 6.
From these results, it was shown that the PowerPAD, when
soldered to the TI recommended board, performed an average of
47% cooler than when not soldered, and 73% cooler than a
standard package on a low-k board.
For the final case, a separate analysis was performed In order to
show the difference In thermal resistance when the standard and the
thermally enhanced peckages are used on the same board. The
results showed that the PowerPAD, when soldered, performed an
average of 44% cooler than the standard package (See Figure 18).

Figure 18. Comparison of BJA for Various Packages

Comparison of Junction-to-Arnbient Thermal
Resistance
a""-PAD._ eTi boMI)
a_AD Il0l_ (TI boMI)

E J _ paduJg8 (Io,v-k-.t)

14p1nTSSOP

48 pin TVSOP

52pk1TQFP

20BplnLCFP

PowerPAO ThermaHy Enhanced Package

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PowerPADTM Thermally Enhanced Package

However, when the PowerPAD Is not soldered to the board, slmMar

to a standard package, the 8..,. Is approximately 3% hotter than a
standard package. This Is due to the location of the lead frame pad
relative to the lead fingers, which Is the strongest conduction path in
a standard package. Since the pad on a standard package lead
frame Is doser to the lead fingers, more heat Is dissipated through
the leads than in the PowerPAD package with ils deep downset pad.

Conclusions
The deep downset pad of a PowerPAD package allows for an
extensive increase In package parformance. Standard packages are
limited by using only the leads to transport a majority of the heat
awey. The addition of a heat sink will Improve standard package
performance, but greatly Increases the cost of a package. The
PowerPAD package Improves performance, but maintains a low
cost. The results of the thermal analysis showed that by soldering
the PowerPAD package directly to a board designed to dissipate
heat, thermal performance increased approximately 44% over the
standard packages used on the same board.

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Figure 18. Comparison of IJJA for Various Packages

Com parison of Junction·to·Am bient Therm al
Resistance

14 pin TSSOP

48 pin TVSOP

52 pin TQFP

208 pin LQFP

However, when the PowerPAD is not soldered to the board, similar
to a standard package, the eJA is approximately 3% hotter than a
standard package. This is due to the location of the lead frame pad
relative to the lead fingers, which is the strongest conduction path in
a standard package. Since the pad on a standard package lead
frame is closer to the lead fingers, more heat is dissipated through
the leads than in the PowerPAD package with its deep downset pad.

Conclusions
The deep downset pad of a PowerPAD package allows for an
extensive increase in package performance. Standard packages are
limited by using only the leads to transport a majority of the heat
away. The addition of a heat sink will improve standard package
performance, but greatly increases the cost of a package. The
PowerPAD package improves performance, but maintains a low
cost. The results of the thermal analysis showed that by soldering
the PowerPAD package directly to a board designed to dissipate
heat, thermal performance increased approximately 44% over the
standard packages used on the same board.

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PowerPADTM Thermally Enhanced Package

Appendix B. Rework Process for Heat Sink TQFP and
TSSOP PowerPAD Packages - from Air-Vac
Engineering

Introduction
The addition of bottom side heat sink attachment has enhanced the
thermal performance of standard surface mounted devices. This has
presented new process requirements to effectively remove, redress,
and replace (rework) these devices due to the hidden and massive
heat sink, coplanarily Issues, and balance of heat to the leads and
heat sink. The following is based on rework of the TQFP1 00 and
TSS0P20124 pin devices.
Figure 19. DRS22C Reworking Station

Equipment
The equipment used was the Alr-Vac Engineering DRS22C hot gas
reflow module. The key requirements for the heat sink applications
include: stable PCB platform with sufficient bottom side preheat,
alignment capabilities, very accurate heat control, and proper nozzle
design.

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PCB support is critical to reduce assembly sagging and to provide a
stable, flat condition throughout the process. The robust convectionbased area heater provides sufficient and accurate bottom side heat
to reduce thermal gradient, minimize local PCB warpage, and
compensate for the heat sink thermal characteristics. The unique
pop-up feature allows visible access to the PCB with multiple easy
position board supports.

Figure 20. Reworking Nozzles of Various Sizes

During removal, alignment, and replacement, the device is held and
positioned by a combination hot gas/hot bar nozzle. Built-in nozzle
tooling positions the device correctly to the heat flow. A vacuum cup
holds the component in place. Hot gas is applied to the top of the
device while hot gas/hot bar heating is applied to the com ponent
leads. The hot bar feature also insures bonding of the fine pitch
leads.

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Figure 21. Nozzle Configuration
HOT CAS

BAR

OR
SOLDER

Profile
The gas temperature, flow, and operator step-by-step instructions
are controlled by an established profile. This allows complete
process repeatability and control with minimal operator Involvement.
Very accurate, low gas flow is required to insure proper tem perature
control of the package and to achieve good solder joint quality.

Removal
The assembly is preheated to 75 ·C. While the assembly continued
to preheat to 100 ·C, the nozzle is preheated. After the preheat
cycle, the nozzle Is lowered and the device is heated until reflow
occurs. Machine settings: TSSOP 20/24 - 220 ·C at 0.39 scfm gas
flow for 50 seconds (preheat) above board level, 220 ·C at 0.39
scfm for 10 seconds. TQFP 100 - 240 ·C at 0.10 scfm for 60
seconds (preheat) above board level, 250 ·C at 0.65 scfm for 15
seconds. The built in vacuum automatically comes on at the end of
the cycle and the nozzle is raised. The time to reach reflow was
approximately 15 seconds. The component is released automatically
allowing the part to fall into an appropriate holder.

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Site Redress
After component removal the site must be cleaned of residual
solder. This may be done by vacuum desoldering or wick. The site is
cleaned with alcohol and lint-free swab. It Is critical that the heat sink
area be flat to allow proper placement on the leads on new device.
Stenciling solder paste is the preferred mathod to apply new solder.
Solder dispensing or reflowing the solder bumps on the pads for the
leads may also be an alternative, but reflow (solid mass) of solder to
the heat sink is not.

Figure 22. Air-Vac Vision System

Alignment
A replacement device is inserted into the gas nozzle and held by
vacuum. The device is raised to allow the optical system to be
utilized. The optical system used for alignment consists of a beamsplitting prism combined with an inspection quality stereo
microscope or cameralvldeo system. the leads of the device are
superimposed over the corresponding land pattern on the board.
This four sided viewing allows quick and accurate operator
alignment.

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PowerPADTM Thermally Enhanced Package

Replacement
Once aligned, the xJy table is locked and the optical system retracts
away from the work area. The preheat cycle is activated. The device
is then lowered to the board. An automatic multi-step process
provides a controlled reflow cycle with repeatable results. Machine
settings for TSSOP 20/24: 160 ·C at 0.39 scfm gas flow for 40
seconds (preheat), 220·C at 0.39 scfm for 60 seconds above board
level, 22O·C at 0.39 scfm for 10 seconds. For TOFP 100: 100·C at
0.78 scfm for 40 seconds (preheat), 240 ·C at 0.10 scfm for 90
seconds above board level, 250·C at 0.65 scfm for 15 seconds (2
stages).

Conclusion
Rework of heat sink devices, TOFP and TSSOP, can be successful
with attention to the additional issues they present. With respect to
proper thermal profiling of the heat sink, die, and lead temperatures,
the correct gas nozzle and profile can be developed to meet the
requirements of the device and assembly. Existing equipment and
nozzle design by Air-Vac can provide the tools and process
knowledge to meat the heat sink TOFP and TSSOP rework
application.

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Appendix C. PowerPAD Process Rework Application Note
from Metcal
The following report references six of Texas Instruments' fine pitch,
surface mount prototype packages (TSOP20, TSOP56, TSOP24,
TOFP1oo, and TOFP64). The shapes and sizes are not new to the
circuit board industry. Normally, I would use Metcal conduction tools
to simply remove and replace these components. However, these
packages are unique because all packages include a 'dye lead' on
the underside of the package. This dye lead cannot be accessed by
contact soldering. Therefore, convection rework methods are
necessary for component placement.
NOTE:
Conduction tools can be used for removal. But, convection
rework techniques are required for placement, and
recommended for removal.)

Removal
Conduction (optional): All packages can be removed with Meteal
conduction tips. Use the following tips:
Component
Metcal Tip Cartridge
OK Nozzle
SMTC-006
SMTC-166
SMTC-006
SMTC-0118
SMTC-112

TSOP20
TSOP56
TSOP24
TOFP100
TOFP64

N-S16
N-TSW32
N-S16
N-P68
N-P20

The dye lead, which is not in contact with the Metcal tip, will easily
reflow as heat passes through the package.

Conduction Procedure
1) lin the tip, contact all perimeter leads simultaneously, and wait
3-5 seconds for the leads to reflow.
2) Uft the package off the board (surface tension will hold it in the
tip cartridge). Dislodge the com ponent from the tip by wiping the
tip cartridge on a damp sponge.

Convection Procedure
1) Flux the leads. Preferably, use a liquid RMNrosin flux. Pre-heat
the board at 100C. Use a convection or IR preheater, like the
SMW-2201 from OK Industries. The settings 2-4 will generally
heat a heavy board to 100· in 60 seconds.

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PowerPADTM Thermally Enhanced Package

2) Remove the component with the OK Industries FCR hot air
system. Use a nozzle that matches the size and shape of the
component (see above). With the preheat still on, heat the top of
the board for 30-45 seconds on a setting of 3-4 (depending on
board thickness and amount of copper in board*).
Since convection Is NECESSARY for placement, convection Is
recommended for removal.

Placement Procedure
1) Pads can be tinned by putting solder peste on the pads and
reflowing with hot air. Simply apply a fine bead of solder paste
(pink nozzle, 24AWG) to the rows of pads. Be sure to apply very
little peste. Excessive paste will cause bridging, especially with
fine pitch components.
2) Once the pads are tinned, apply gel flux (or liquid flux) to the
pads. RMA flux is preferable. Be sure to apply gel flux to the dye
pad as well. It is important that your pads not be OVER tinned. If
too much solder has formed on the dye pad, the component will
sit above the perimeter leads, causing co-planarity problems.
The gel flux is tacky and helps with manual placement. The
joints require very little solder, so stenciling is not necessary.
The pads are so thin that a minimal amount of solder is needed
to form a good joint. Use a hot air nozzle for the FCR system.
Pre-heat the board and (setting 3-5). Use low air flow (5-10
liters/minute) and topside heat (setting 3-4) for about 30-45
seconds*.
NOTES:

The quality of the dye lead's solder joint cannot be visually
inspected. An X-ray machine, cross sectioning, or
electrical testing will be required.
The vias on the test board are not solder masked very well
which causes some bridging and solder wicking.
*Specific board and component temperatures will vary from board to board and from nozzle to
nozzle. Larger nozzles require a higher setting because the heat must travel farther _ay from the
heat source. There will be a slight convection cooling effect from pushing hot air through long flutes,
and depending on how wide the nozzle is. However, as a rule, keep the board temperature at 100
'C (as thermocoupled from the TOP). You can regulate the board temperature by setting the
temperature knob on the bottom side pre-heater. Apply a HIGHER topside heat from the FCR
heating head. As a rule, use a maximum of 200-210'C for a short pesk period (10 seconds). look
for the flux to bum off. For board profiling purposes, you can visually inspect the condition of the
solder joints during the removal process. Note the time allotted for reflow and sat the system to Auto
Remove or Auto Place at the same lime designation for good repeatability. Be sure not to overheat
the joints. Excesslva heat can cause board delamination and discoloration. Alignment will 'seWcorrect' once all the solder has ref\owed. Tap board lightly. Remove any solder bridges with solder
braid. Also, limit the board's heating cycles to a minimum. Excessive heat shock may warp the
board or cause cracking in the solder joints.

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Reducing and Eliminating the
Class-D Output Filter
Application Report

SLOA023
August 1999

~TEXAS

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Printed on Recycled Paper

5-85

IMPORTANT NonCE

Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue
any product or service without notice, and advise customers to obtain the latest version of relevant information
to verify, before placing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those
pertaining to warranty, patent infringement, and limitation of liability.
TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with TI's standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily
performed, except those mandated by govemment requirements.
CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
DEATH, PERSONAL INJURY, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE ("CRITICAL
APPLICATIONS"). TI SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, AUTHORIZED, OR
WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT DEVICES OR SYSTEMS OR OTHER
CRITICAL APPLICATIONS. INCLUSION OFTI PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO
BE FULLY AT THE CUSTOMER'S RISK.
In order to minimize risks associated with the customer's applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. TI's publication of information regarding any third
party's products or services does not constitute TI's approval, warranty or endorsement thereof.

Copyright © 1999, Texas Instruments Incorporated

5-86

Contents
1 Introduction ••••••.••.•••••.•••••..•.•..•..•..•••.•..•.••••.•••.••••••.•••.••............... 5-89
2 Second-Order Butterworth Low-Pass Filter •.•.•......•...•..••.....•...•..•...•..•...•.••.•.. 5-90
3 Half Filter .......•••....•••••••..•.••...••.••.••••••....••••.••••••••••••••••••••..••.••.•••• 5-91
4 No Filter ..••••••••••..•.••....•..........•.....•..•....•••••.••.••..••..••.••.••••••.••.•••• 5-92
5 Speaker Selection •.••..•.••.••••..••....•••.•••.•••.••••••.•••.••••••..••.••.•.••••..••.•..• 5-93
5.1 Class-D With Full Filter and Half Filter .................................................... 5-93
5.1 .1 Zobel Networks Reduce Peaking ................................................. 5-93
5.2 Class-D Without Filter .................................................................. 5-94
5.2.1 High-Inductance Speakers ...................................................... 5-94
5.2.2 Speakers with Slightly Higher Power Rating ....................................... 5-97
6 Speaker Selection ••••....•.....••..•.....••..••.•••.....••••••••...••..•..••••.••.••.•..•..• 5-98
7 Quiescent Current •••••..••...•••..••....••...•••.•......••..•••...••..•••••••..•••..•••..• 5-100
8 Fidelity .................................................................................... 5-101
8.1 Total Harmonic Distortion Plus Noise (THD+N) ........................................... 5-101
8.2 Intermodulation Distortion (IMD) ........................................................ 5-103
9 Electromagnetic Interference (EMI) .•••...••....••••......•.••...•.•••...•••••••.••••.••..••• 5-105
9.1 E and H Field Measurements .......................................................... 5-105
9.2 EMI Measurement Conclusions ......................................................... 5-108
9.3 Reducing EMI ........................................................................ 5-109
10 Filter Selection from System Requirements ....••••...•.•••..•••..•.•••...••••..••••••.•.••. 5-113
10.1 No Output Filter ..................................................................... 5-113
10.2 Half Filter ........................................................................... 5-114
10.3 Full Filter ........................................................................... 5-114
11

Conclusion .••.••.••••••.•••••••..••...•••....••.•....•.••.•••..•...•.•.••••.••••••••.•.•• 5-115

12 References •.•.••...•.••••••••••..•••.•••..•.•••••...••.•..•••..•...•...•.••.••••.•••.•••• 5-115

Reducing and Eliminating the Class-D Output Filter

5-87

Figures

List of Figures
1 Full Second-Order Butterworth Filter ............................................................. 5-90
2 Half Filter ..................................................................................... 5-91
3 Class-D Amplifier With Zobel Network ............................................................ 5-93
4 TI Speaker Impedance vs Frequency ............................................................ 5-95
5 TI Speaker Phase vs. Frequency ................................................................ 5-95
6 Total Harmonic Distortion Plus Noise vs Output Voltage ........................................... 5-102
7 Total Harmonic Distortion Plus NOise vs Frequency ............................................... 5-102
8 SMPTE Intermodulation Distortion vs Input Voltage ............................................... 5-102
9 CCIF Intermodulation Distortion vs Difference Frequency .......................................... 5-104
10 E Field Measured 1h Inch Above Speaker Wire .................................................. 5-106
11 H Field Measured 1h Inch Above Speaker Wire .................................................. 5-106
12 E Field Measured 1h Inch Above Class-D Output Traces .......................................... 5-107
13 H Field Measured 1h Inch Above Class-D Output Traces .......................................... 5-108
14 Shielded Twisted Pair Speaker Connection ..................................................... 5-110
15 Standard Speaker Wire and Shielded Twisted Pair E Field vs lime ................................ 5-111
16 Standard Speaker Wire and Shielded Twisted Pair H Field vs Time ................................ 5-111

List of Tables
Additional Quiescent Current per Channel from Switching Loss in Speaker Without Filter .............. 5-96
2 Quiescent Current for Various Filter Applications Using the TPA005D02 and the TPA0102 ............. 5-100
3 Performance Ranking of Full Filter, Half Filter, and No Filter Applications ............................ 5-113

5-88

SLOA023

Reducing and Eliminating the Class-D Output Filter
Michael D. Score
ABSTRACT
This application report investigates reducing and eliminating the LC output filter
traditionally used in class-D audio power amplifier applications. The filter can be
completely eliminated if the designer is using a predominantly inductive speaker;
however, the supply current and the EMI are higher than if using the full second-order
Butterworth low-pass filter. The designer can use half of the components in the originally
recommended second-order Butterworth low-pass filter to reduce the supply current, but
the EMI is still higher than that of the full filter. The half and no filter class-D applications
outperform the full second-order Butterworth filter in total harmonic distortion plus noise
(THD+N) and intermodulation distortion (IMD). This document shows speaker
requirements with and without a filter, fidelity and electromagnetic Interference (EMI)
results, and indicates what type of filter fits various system requirements.

1 Introduction
A properly designed class-D output filter provides many advantages by limiting
supply current, minimizing electromagnetic interference (EMI), and protecting
the speaker from switching waveforms. However, it also significantly increases
the total solution cost. The current recommended second-order output filter for
the TPA005D02 is 30% of the audio power amplifier (APA) solution cost. This
application report discusses the recommended second-order Butterworth filter
as well as two reduced filtering techniques, each providing a different
price/performance node. The first alternative to the Butterworth filter reduces the
output filter by half and the second option completely eliminates the filter.
The total harmonic distortion plus noise (THD+N) and intermodulation distortion
(IMD) of the class-D amplifier with full filter, half filter, and no filter were measured
using a Texas Instruments TPA005D02 Class-D APA. Near-field EMI was
measured and methods to reduce EMI are suggested for each application. Filter
selections were then made based on system requirements.
This document gives speaker and filter component recommendations for each
filter application.

5-89

Second-Order Bunerworth Low-Pass Filter

2

Second-Order Butterworth Low-Pass Filter
The second-order Butterworth low-pass filter is the most common filter used in
class-O amplifier applications. The second-order Butterworth low-pass filter as
shown in Figure 1 uses two inductors and three capacitors for a bridged-tied load
(BTL) output [1].

Figure 1. Full Second-Order Butterworth Filter

The primary purpose of this filter is to act as an inductor to keep the output current
constant while the voltage is switching. If the amplifier outputs do not see an
inductive load at the switching frequency, the supply current will increase until the
device becomes unstable. Higher inductance at the output yields lower quiescent
current (supply current with no input), because it limits the amount of output ripple
current.
The filter also protects the speaker by attenuating the ultrasonic switching signal.
Inductors L1 and L2, and capacitor C1 form a differential filter that attenuates the
signal with a slope of 40 dB per decade. The majority of the switching current
flows through C1, C2, and C3, leaving very little current to be dissipated by the
speaker. The filter also greatly reduces EMI, which is discussed in a subsequent
section.

'5-90

SLOA023

Half Filter

3 Half Filter
The half filter, as shown in Figure 2, eliminates one of the inductors and the two
capacitors to ground from the full filter.
L

~~--------lr'------~

r,----LLJ

Figure 2. Half Filter
For the cut-off frequency to remain unchanged, the value of the inductor is
doubled while the value of the capacitor across the load stays the same. The
capacitors to ground are removed to prevent one of the amplifier outputs from
seeing a capacitive load, which would greatly increase the supply current. This
filter is still inductive at the switching frequency because the capacitor looks like
a short at that frequency.
Aside from the primary advantage of reduced system cost, the half filter also
decreases the quiescent current. In the case of the full filter, part of the switching
current is shunted to ground through one of the capacitors. In the half filter, the
absence of a capacitor to ground eliminates this waste. Furthermore, each output
sees the full inductance value, which effectively reduces the rate of change in the
inductor current, providing less power loss in the filter. Although this filter
attenuates the differential signal, which reduces the magnetic field radiation, it
does not attenuate the common mode signal, which causes the electric field
radiation. Sources of EMI and methods to reduce EMI are covered in Section 9.

Reducing and Eliminating the Class-D Output Filter

5-91

No Filter

4 No Filter
The filter can be completely eliminated if the speaker is inductive at the switching
frequency. For example, the filter can be eliminated if the class-O audio power
amplifier is driving a midrange speaker with a highly inductive voice coil, but
cannot be eliminated if it is driving a tweeter or piezo electric speaker, neither of
which are highly inductive. The class-O amplifier outputs a pulse-width
modulated (PWM) square wave, which is the sum of the switching waveform and
the amplified input audio signal. The human ear acts as a band-pass filter such
that only the frequencies between approximately 20 Hz and 20 kHz are passed.
The switching frequency components are much greater than 20 kHz, so the only
signal heard is the amplified input audio signal.
The main drawback to eliminating the filter is that the power from the switching
waveform is dissipated in the speaker, which leads to a higher quiescent current,
IOO(q). The speaker is both resistive and reactive, whereas an LC filter is almost
purely reactive. A more inductive speaker yields lower quiescent current, so a
multilayer voice coil speaker is ideal in this application.
The switching waveform, driven directly into the speaker, may damage the
speaker. The rail-to-rail square wave driving the speaker when power is applied
to the amplifier is the first concern. With a 250-kHz switching frequency, however,
this is not as significant because the speaker cone movement is proportional to
1/f2 for frequencies beyond the audio band. Therefore, the amount of cone
movement at the switching frequency is insignificant [2]. However, damage could
occur to the speaker if the voice coil is not designed to handle the additional
power. Section 5 focuses on selecting the speaker and includes a derivation for
choosing the power requirements of the speaker when not using an output filter.
Eliminating the filter also causes the amplifier to radiate EMI from the wires
connecting the amplifier to the speaker. Therefore, the filterless application is not
recommended for EMI sensitive applications. Methods of reducing EMI are
discussed in Section 9.

5-92

SLOA023

Speaker Selection

5 Speaker Selection
5.1

Class-O With Full Filter and Half Filter
Selecting an appropriate speaker for a half-filter or full-filter class-D application
is approximately the same as specifying a speaker for a class-AS application.
First, the speaker should be efficient, or it should provide better than average
sound pressure level (SPL) output for a given power input. Second, the speaker
must also have a good frequency response, meaning a relatively constant SPL
across a wide frequency range for a given input power.
A speaker should have a low inductance voice coil if designing with a filter, as the
inductance causes a peak to appear in the output at the corner frequency of the
filter. Peaking is not a significant problem in class-D applications though, because
the corner frequency of the filter is set outside the audible frequencies. The
class-D output filter should have a corner frequency of 25 kHz or higher, so the
peaking may slightly affect the upper frequencies of the audio band. However,
this peaking should be so small that it has an insignificant effect on the sound
quality.

5.1.1 Zobel Networks Reduce Peaking
If the peaking does cause problems in a given system, a simple RC matching
network, also called a Zobel network, may be placed in parallel with the speaker,
as shown in Figure 3.

Class-D
Audio
Power
Amplifier

Filter

Figure 3. Class-D Amplifier With Zobel Network
The resistor and capacitor act to dampen the reactance of the load. The
equations for the components of the Zobel network are shown in equations 1 and
2. RL is the DC resistance of the speaker, and can be measured with an
ohmmeter. Le is the electrical inductance at DC, and is usually given as a speaker
parameter. The power rating of the resistor and capacitor of the Zobel network
are dependent upon the selected component values and must be calculated. The
power rating of the resistor will be high for many applications, making this solution
impractical for many systems in which cost and size are important.
Rz == 1.25

Cz =

LE

R2

X

RL

(1)
(2)

L

Reducing and Eliminating the Class-O Output Filter

5-93

Speaker Selection

5.2 Class-D Without Filter
The major difference in selecting a speaker for a class-D amplifier without a filter
is that the speaker must have a high inductance. Furthermore, the speaker power
rating must be slightly increased to account for the switching waveform being
diSSipated by the speaker instead of by the filter.

5.2.1

High-Inductance Speakers
The filterless class-D application requires a speaker with a high inductance to
keep the output current relatively constant while the output voltage is switching.
As a result, the filterless approach may be impractical for use with a tweeter or
a piezo electric speaker, both of which typically have small inductances. Without
the filter, the peaking problem with the full and half filter application disappears
because there is no filter to form a resonant circuit.
The additional quiescent current due to switching waveform power dissipation in
the speaker can be calculated by first thinking of the speaker as a complex load.
The switching current diSSipated in the speaker can be calculated if the
impedance and phase of the speaker is known for frequencies greater than the
switching frequency. The magnitude and phase of the impedance of the speaker
may be measured with a network analyzer from the switching frequency and
higher to get an exact measurement on the switching loss in the speaker.
2

~ vsw(n x fsw) x COS(CPSPKA (n Xfsw))

POlS =

I

)
IZSPKA (
n x fsw I

n-l

(3)

The Fourier series needs to be calculated for the switching waveform that is being
applied to the speaker. This is not as difficult with the TPA005D02, which has the
standard modulation scheme, because the switching waveform voltage, Vsw, is
a square wave, which is composed of the sum of many sine waves with
frequencies of the odd harmonics of the switching frequency. The RMS value of
the harmonics are shown in equation 4. The impedance and phase must then be
calculated at each odd harmonic of the switching frequency.
v SW ( n x fSW )

5-94

SLOA023

=

0.707 x Voo
n
for n

=

1, 3, 5, 7, 9, 11, ...

(4)

Speaker Selection

The impedance and phase of the speaker that Texas Instruments provides with
the TI Plug-N-Play Audio Evaluation Platform can be seen in Figure 4 and
Figure 5.

,

600

500

I \

400

c:

300

I

200

8c
01

I.§

V

100

i

30

I

20

./

I

VI

!:i

I

10

v

II

o
10

I

1k

100

J

/

10k

100k

1M

10M

f - Frequency - Hz

Figure 4. TI Speaker Impedance vs Frequency

60
L

40

I

i

i

,

r-...

\
,..,..

20

o

\

,./

"""" ~
\

-20

\

-40

-eo

\
10

100

1k

10k

100k

1M

10M

f - Frequency - Hz

Figure 5. TI Speaker Phase vs Frequency
Reducing and Eliminating the Class-D Output Filter

5-95

Speaker Selection

After measuring and calculating the value of the components at the harmonics,
equation S can be used to calculate the added current drawn from the supply. The
constant 0.S8 is required in finding the RMS current from the peak current of a
triangle wave.

~

L\IOO(q) =

I

n-l

0.58 x Voo x COS( ~ VR.1c::n:

CJ)

0

0

0

'" '"

L

Audio
Power 0
Amps 0

000
0

°0
0

L

8
0
0
0
0

r------

~
°

S

0
0

0
0
-I

m

~

0
0

I

~

m

-c:JC

DC

C»

Power
In/Out

:0

g-1iiI'IQl ar

8--Rt
.

z"'O

0'"

a

0
0
0

I
-01 I '-2en" or
c'2
sa
MOde I
Mule

'-

dl

:I: PolarIIy

000
C

Gl-l

Speaker
Output

000

01

Plug-N-Play Audio Amplifier
Evaluation Platform
SLOP097 Rev. C.1

8

0

~o

0
~g 00
..,a. 00 00
ci"
.,§o - 8
0
0
0
0
0

om

02

000

0
0
0
0
0

",Sc:..

C::

"' ...
[JJ
C

en:I:en
g"'U w
11
CD

La
H'I

"tJ
.."

+

c...
CO

I})

"''ii

s.~

Oir
O~

R3 +

HPOut c...

00

"'IIIIIIIIIIID _ _ _ R5
R4---

Inuoducuon

6-15

Description

The audio EVM sockets are arranged into two stages on the platform (Figure
1-2): an input signal conditioning stage (socket U1) and an output power
amplifier stage (sockets U2 through U5). The signal conditioning EVM can
include such functions as volume and tone controls as well as the mixing of
several sources, and can be bypassed with a switch on the platform. The
output amplifier stage can be populated with a wide variety of EVMs, including
both single-channel and stereo units, and is intended to drive speakers and
headphones.

Figure 1-2. Functional Block Diagram

F;~~~~-Zl-:

=R--....------R
•

.... ..... • ".... - ,;," .. "_1

Speaker

Output
L
Full-Wave
Bridge
Rectifier

JP2

JP1

~~----------

__D

U3lU4

orU2

:;~ 'T"~:" :~;.- ;;.:J·;';J-'R'"'----,I--.. . . .

...-_R.

.A,

',S3
, '"

f~;:·~;~';~';·~~~:i'h;.~:~:.:

On

Headphone

'---...;-....-L··

Output

US

Signals are input through either a pair of RCA phono jacks (left and right
channels) or a miniature stereo phone jack. These inputs are grounded when
the jacks are not in use. Signal conditioning EVMs may have additional input
connectors, as in the case of the Microphone Mixer EVM (SLOP107), which
has a microphone input jack mounted on its circuit board.
The platform includes a pair of sockets for single-channel power amplifiers (U3
and U4) and a socket for a stereo power amplifier (U2). These sockets physically overlap each other such that either one or both mono amplifiers can be
installed, OR, a single stereo amplifier can be installed - but not any combination of stereo with mono amplifiers. Outputto speakers is through a pair of RCA
phono jacks and compression connectors for use with stripped speaker wire.
Socket U5 is typically for a stereo headphone amplifier EVM. A miniature
stereo headphone jack is capacitively coupled to either the headphone
amplifier outputs or the power amplifier outputs as selected by a switch.
The platform Vee supply can be provided by a wide variety of sources,
including an on-board 9-V battery for low-power or short-duration projects and
unregulated external AC or DC between 5 V and 15 V for other applications.
ForTI audio EVMs that require a regulated 3.3-V or 5-V Voo supply, a voltage
regulator EVM can be installed on the platform (U6), or external regulated Voo
power can be applied to a connector on the platform.

6-16

Introduction

Chapter 2

Quick Start
This chapter contains a quick-start list that explains how to configure the
platform, connect power, connect the inputs and outputs, and power up the
system.

Topic
2,,2'
12
.

Page
Precautions

. Quick Start List

. :::: I

6-17

Precautions

2.1

Precautions

Figure 2-1. Quick Start Map

<0

:lJ

en
C

""

!:( m
JJ

I

Power
Input

00

@L

'g- IiiI
IiiI dl

8----

.

000
C

en

Gl-l

°
6-18

z-C
0 .....

@

---- •

000

I\)

0
0
0

"::lE0

0
0
0

a

Quick Start

Quick Start List

2.2 Quick Start List
The following steps can be followed to quickly prepare the TI Plug-N-Play
Audio Amplifier Evaluation Platform and EVMs for use. Numbered callouts for
selected steps are shown in Figure 2-1 and details for each step appear in
Chapter 3.

o

Configure the platform
1) Ensure that all external power sources are set to OFFand that the platform
power switch 81 is set to OFF; set gain controls to minimum
2)

Select the TI audio evaluation modules to be used

3)

Install the modules on the platform in the appropriate sockets

4) Use switch 82 to select or bypass the signal conditioning EVM (U1)
5)

If the headphone jack (J1 0) output will be used, set source switch 83 to
U5 or U2-U4 according to which sockets have power amplifiers installed

6) Consult the User's Guide for the power amplifier installed in U5 (if any) and
set control signal Polarity jumper JP7 to either HI or Lo

7) Consult the User's Guide for the power amplifiers installed in U2-U4 (if
any) and set control signal Polarity jumper JP8 to either HI or Lo
8) Consult the User's Guide for the power amplifiers installed in U2-U4 (if
any) and set jumper JP6 to select either the Mute or Mode control input

o

Connect power supplies
9) Consult the User's Guides for the modules installed and select external
power supplies that will provide a voltage appropriate for the modules
installed (platform Vee must be within the range of 5.5 V to 15 V, or 5.5 V
to 12 V with a SLVP097 regulator module installed in U6, for example)
10) If any module installed on the platform requires a regulated VOO of 3.3 V
or 5 V for operation, install a SLVP097 regulator EVM (or equivalent) in U6
or connect an external regulated power supply adjusted to the correct
voltage to screw terminals J6, taking care to observe marked polarity
11) Connect power to, and jumper ONE of the following Vee power inputs:
a) Connect an external DC power supply to screw terminals J1, taking
care to observe marked polarity, and jumper JP1

o

b)

Plug a coaxial power connector (AC or DC) into J2 and jumper JP2

c)

Install a 9-V battery into 81 and jumper JP3

Connect Inputs and outputs
12) Connect the audio source to left and right RCA phono jacks J3 and J5 or
stereo miniature phone jack J4
13) Connect 4-n - 8-n speakers to left and right RCA jacks J7 and J9 or to
stripped wire connectors J8, or plug headphones into J10

o

Powerup
14) Verify correct voltage input polarity and set external power supplies to ON,
then set platform power switch 81 to ON
LED1 should light indicating the presence of Vee, LED2 should light indicating
the presence of Voo (if used), and the evaluation modules installed on the
platform should begin operation.
15) Adjust signal source levels and EVM gain controls as needed

Quick Start

6-19

6-20

Quick Start

Chapter 3

Details
This chapter provides details on the steps in the Quick-Start List and additional
information on the TI Plug-N-Play Audio Amplifier Evaluation Platform.

Topic
3.1

Page
Precautions. • .. • • • • . • • • • • • • • • . • • • .. • • . • . • • • • .. . . . .. . • .. • ... s.,.22

,: 3.2.. 'Conflguratlon ••••••• ~ ......... , .................... , •••'. • • •• $-23
3.2.1 TI Audio EVMs , '," .......................... ,...•• \. '..... 6,-,23
3.2.2 Installing and FjemO\ling EVrvI Boards .........•...•• , ... ~ •. ,s.,.23
3.2.3 Signal Routing ..•. '.•....•. ,: •....••...•.••..... ; • ; \'~ . ,; ;s.,.24
3.2.4 Muting and Mode ..••.....•••...•... ; .•.•.••...•..... , .';, s.,.25,

POwer , ............................................... ~ •••• '" s.,.27
3.3.1 Platform Power Distribution .......................•.. i . • , s.,.2J
3.3.2 Platform Power Protection ............ , .............. ,.... s.,.27
3.3.3 Platform Power Inputs ............................. ;..... s.,.28
Inputs and Outputs. • • • .. • • . . • . • • . • . • . . • . • . . . • . . • . . . • • • • . • • •• 6-31
3.4.1 Inputs .............................................. ;.. s.,.s1
3.4.2 Outputs. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . • . . . . . . .. s.,.s2

:U Troubleshooting ......... , • • • • • • • • • • • • • • • .. • • • • .. • • • • . .. • • •• 6-33

.3;6 ';Parts list. • • • .. • • • • . • • • • • • • .. • • • • .. • • • • • • • • • • •• .. • • • • • .. •• •• 6-34
~'

Platform EVM Socket Pinouts ................................ 6-35

s.,.21

Precautions

3.1

Precautions

..

Figure 3-1. The Platform

°
~

~gl~I~I~_C10·

... .---

< 0
+3"8~m

Power
Input

L

< 'i::> III
8~§ III

VRh

---.------+ +
R

Stel'el) Poltler: R

1---+-------+ - J7,J8,J9

AmPlifier:

I

Speaker
Outputs

.Of·.

114 $nlflor U3
: Mono pOWtt,
I

!J-!
~ I '.~~......
. ~-.~.-.--'

._~

I

J3, J4, J5
Audio
Input

I
U1
I
.. 'SinriaJ
I " .
I

I
I
I

CQ.ncfl1ionIl'i9'.I

. : L

L

~._,;;~"!!f~~!_ ;;.1--'--+-U-2---U-4---+ +

I

: S2

_____

R

~.

.us .......

~+

R

J10
D - Headphone
•.. ,:S~ '.'
:, S3 G:FC
Output
~'~~~!~:t~t---t:

"..____,' HflSdphori!l,'
On

us
Switch S3 is the source select for the stereo headphone output jack, J 10. The
headphone jack is capacitively coupled and can output either the Signal from
the headphone amplifier in socket U5, or the output Signal from the power
amplifier(s) installed in socket U2 or U3 and/or U4, as determined by the
setting of S3.

6-24

Details

Configuration

3.2.4

Muting and Mode
Many TI audio power amplifier EVMs have control inputs that mute the output
and/or change the operating mode (from bridged to single-ended output, for
example) in response to a signal applied to a control input. The typical
application, as often found in notebook computers, portable audio products,
and such, would have the internal, speakers mute when headphones are
plugged into the headphone jack, .or have internal speakers mute when
external speakers are connected.
In applications using separate speaker and headphone amplifiers, the one not
being used can be shut down (muted) to conserve power. In applications that
use a single power amplifier to run either the speakers or the headphones, or
either the internal speakers or the external speakers, often the amplifier must
switch its output mode to single-ended to be able to cope with the 3-wire
headphone or extemal speaker connector that returns the signal to ground.
The TI Plug-N-Play Audio Amplifier Evaluation Platform has been designed to
provide complete flexibility in selecting control signal polarity and functionality
for amplifier muting and mode select.

3.2.4.1

Headphone Jack Control Signals
The platform headphone jack (J10) contains an internal switch that changes
the state of a pair of control lines when a headphone plug is inserted. Each
control line is pulled down by a 1-1<0 resistor to ground (R4 and RS). The switch
in the headphone jack pulls one line or the other up to Voo through a 240 n
resistor (R3) depending on whether a headphone plug is inserted in J1 0 or not
(Figure 3-3).

Figure 3--3. Mute/Mode Control
VDD

Mute

:

U5

:

:

Amplifier

:

r-------------------===-~ HNdphone •
JP7 PolarIty JP8

J10

Headphone
Jack

JP6
o-t...:M",od=e-...,: 0+-+0
•
Power
•
.:,...I-t-----...-----+iJ-f---::-::----t-D
O+--:MC:-ute--:--'~~ _ ~~~I~~ _

Lo

U2, US, U4 -:

!

T

R4
1 kn

A 3-pin jumper header (JP7), functioning as a SPOT switch, selects the control
signal polarity by connecting either the active-low or the active-high line from
the headphone jack to the mute control input of the headphone amplifier
socket, US.
For the power amplifiers, sockets U2 - U4, a second three-pin jumper header
(JP8) selects the control signal polarity by connecting either the active-low or
the active-high line from the headphone jack to jumper JP6. JP6 connects the
control signal from the headphone jack to either the mute or the mode control
input of the power amplifier sockets.

Details

6-25

Configuration
3.2.4.2 Muting Polarity Select for Headphone Amplifier In US (JP7)
Jumper JP7 as indicated in the User's Guide for the amplifier installed in U5,
or:

o

To mute EVMs that are being used as headphone amplifiers in U5 when
the plug is removed from the headphone jack, jumper JP7 as follows:
•

If the EVM mutes on a low control signal, jumper JP7 to HI

•

If the EVM mutes on a high control signal, jumper JP7 to Lo

3.2.4.3 MuteiMode Select for Power Amplifiers in U3/U4 or U2 (JP6)
Jumper JP6 as indicated in the User's Guide for the power amplifiers installed
in U2 or in U3IU4, or:

o
o

To change the mode (from BTL to SE, for example) of the power amplifiers
installed in U3IU4 or U2 when a plug is inserted in the headphone jack,
jumper JP6 to Mode
To mute the power amplifiers installed in U3/U4 or U2 when a plug is
inserted in the headphone jack, jumper JP6 to Mute

3.2.4.4 Mute/Mode Polarity Select for Power Amplifiers In U3/U4 or U2 (JPB)
Jumper JP8 as indicated in the User's Guide for the power amplifiers installed
in U2 or in U3/U4, or:

o

To mute or change the mode of the power amplifiers installed in U3/U4 or
U2 when a plug is inserted in the headphone jack, jumper JP8 as follows:
•

If the power amplifiers mute or change to the desired mode on a low
control signal, jumper JP8 to Lo

•

If the power amplifiers mute or change to the desired mode on a high
control signal, jumper JP8 to Hi

3.2.4.S Mute/Mode Jumper Select Table
Table 3-1 shows the relationship between the control line polarity select
jumpers (JP7 and JP8), the Mute/Mode select jumper (JP6), and the
headphone plug for amplifier EVMs with active-high control inputs.

Table 4-1. Mute/ModelPolarlty Jumper Select Table
POWER AMPLIFIERS
JP6

JP8
Lo

Mute
Hi
Lo
Mode
Hi

6-26

U2-U4

HEADPHONE
PLUG

HEADPHONE AMPLIFIER
US

Active

Present

Active

Mute

Not present

Mute

Mute

Present

Mute

Active

Not present

Active

Mode A

Present

ModeB

Not present

ModeB

Present

Mode A

Not present

JP7
Lo
Hi

Details

Power

3.3

Power
TI audio modules installed in the TI Plug-N-Play Audio Amplifier Evaluation
Platform operate from either an unregulated Vee supply or a regulated Voo
supply. The platform can be powered from an on-board battery or from any of
several different external sources.

3.3.1

Platform Power Distribution
The platform is equipped with a number of connectors for power input, and a
Vee bus and a Voo line for on-board power distribution. The Vee bus uses
jumper block JP1 OR JP2 OR JP3 to connect it to the desired power input
connector. Only ONE of these jumpers should be installed at anyone time.
On-board switch S1 applies Vee power to the modules installed on the
platform. Note that S1 also controls Voo power only when Voo is supplied by
a power supply/Voltage regulator module plugged into platform socket U6, and
not when Voo power is supplied from an external source at screw terminals
J6. LE01 and LE02 indicate the presence of power on the Vee bus and the
Voo line, respectively (Figure 3-4).

Figure 3-4. Platform Power Distribution

Vcc

ICC

S1

IDD
LED1

VR1

C1

470~F

LED2

~

R2

Jumper JP4 is in series with the Vee bus and allows easy monitoring of module
Vee current consumption (ICc). JP5 is in series with the Voo line for 100
measurement. Both current monitoring points are on the load side of the
indicator LEOs, so their current consumption will not be part of the
measurement.

3.3.2

Platform Power Protection
The platform Vee bus and the Voo line are protected against both excessive
voltage levels and reverse power polarity by zener diodes and fuses
connected to form crowbar circuits.
A zener diode is connected backwards between the Vee bus and ground so
that it is reverse-biased. If the input voltage exceeds the zener breakover
voltage, the diode suddenly conducts heavily, forming a low-impedance path
to ground. The resulting high current opens the fuse, removing the voltage.

Details

6-27

Power
If a reverse-polarity voltage is input, the zener, being forward biased, conducts
immediately, and again the fuse opens. A similar circuit protects the V DO line.
Note that the vee bus protection components are ahead of the platform power
switch. And since there is no power switch for the Voo line, both protection
circuits will respond to reverse power polarity and overvoltage conditions at
the moment they are applied to the platform power input connectors. Power
polarity and voltage levels must be set and verified before external power is
applied to the platform.
Correct polarity and maximum voltage levels should always be strictly
observed because not only is the operation of the crowbar circuit always
destructive to some degree (at a minimum, the opened fuse must be
unsoldered and replaced), there is always the chance for damage to the
platform, the installed modules, and/or the external power source before the
fuse opens.
Damage to the protection circuit and/or the platform (beyond an open fuse)
can occur if the external power supply is unable to provide at least 3 amps of
current to ensure the fuse opens quickly. Lower currents can cause failure of
the zener diode and possibly damage to the platform PCB traces from
overheating.

3.3.3 Platform Power Inputs
The evaluation modules installed on the platform can be powered by a wide
variety of Vee sources including:
DOn-board 9-V battery

o
o

Unregulated external DC at screw terminals J1
Unregulated external AC or DC at coaxial power connector J2

And for those TI audio EVMs that require a regulated Voo supply:

o
o

Regulated DC from on-board power supply/regulator (socket U6)
Regulated external DC at screw terminals J6

Selecting the appropriate power source may depend on the requirements of
the various modules in the audio system assembled on the platform, or simply
on what is available (as long as platform and EVM requirements are met).

3.3.3.1

Power Requirements
Platform Vee voltage limits are governed by the lowest level that will operate
all of the installed modules and the highest level that the modules (or the
platform overvoltage protection circuit) will tolerate. In general, however, the
Vee input voltage should be in the range of:

6-28

•

Approximately 3.3 V to a maximum of 15 V

•

Approximately 5.5 V to a maximum of 12 V if a SLVP097 power
supplylvoltage regulator module is installed in U6

Details

Power
Some TI audio EVMs require a regulated Voo supply (3.3 V or 5 V typical) for
operation. This can be provided by a power supplylvoltage regulator EVM
installed in platform socket US (runs off of the platform Vee bus) or by an
external regulated supply. If an external Voo source is used, depending on the
EVM reqUirements, Voo should be:
•

3.3.3.2

3.3 V or 5 V, and must not exceed S V

On-Board 9-V Battery
Many low-power portable and battery-powered audio systems can be
modeled on the platform with TI audio EVMs. It may make sense, then, to
power these system mOdels on the platform using an on-board battery. The
platform is equipped with a snap-in battery holder for a common 9-V battery
and jumper JP3 connects the battery to the Vee bus, which routes the battery
voltage to the various EVM sockets.
Since the Vee bus also supplies the on-board power supplylvoltage regulator
socket, the battery voltage can be input voltage for a power supply/regulator
EVM plugged into US. The regulator EVM then supplies regulated voltage to
the Voo line for use by those EVMs that require regulated Voo power.
For high-power audio system evaluation and demonstration, one of the other
platform power supply options should be selected.

3.3.3.3

Unregulated External DC at Screw Terminals J1
Unregulated DC voltage from a bench-type supply or any other source of DC
power within the required voltage range can be connected to screw terminals
J1 for Vee power. Jumper JP1 connects J1 to the Vee bus for distribution.
Voltage applied to screw terminals J1 MUST be of the correct polarity and
MUST NOTexceed 15 V or the power protection circuit on the Vee bus will trip.

3.3.3.4

Unregulated External AC or DC at Coaxial Power Connector J2
The coaxial power jack, J2, matches a large number of the typical wall-cubetype power transformers/power supplies. Although the jack is of a standard
size (5.5 mm 0.0. x 2.1 mm 1.0.), there does not seem to be any standard for
voltage polarity or power type (AC or DC) among wall-cubes and other power
sources using a coaxial power plug. To ensure the widest possibility
compatibility, the platform is equipped with a full-wave bridge rectifier between
the coaxial connector and the Vee bus to allow DC voltage of either polarity,
or AC voltage to be input through J2. Jumper JP2 connects J2 to the Vee bus
for distribution.
The bridge rectifier eliminates the need to determine the plug polarity for input
voltage at J2 and rectifies AC voltage applied to J2 into DC before it is
connected to the Vee bus. An on-board filter capacitor on the bus smooths the
rectified AC.
With DC voltage applied to J2, the bridge rectifier introduces a voltage drop
of approximately 1.4 V (two diode forward-drops). This drop must betaken into
account if the DC voltage applied to J2 is at or near the minimum required for
operating a module installed on the platform, and the external voltage supply
adjusted accordingly.

Details

6-29

Power

With an AC voltage applied to J2, Vee bus voltage depends on several factors,
including the load on the bus. As a general rule for typical AC voltage inputs,
however, Vee bus voltage will be approximately the peak value of the applied
ACvoltage.
The bridge rectifier also causes the platform ground bus to be approximately
0.7 V above the ground of other equipment that might be operated by the same
external power supply. Platform Vee and EVM voltage measurements should
be referenced to the platform ground bus (test point TP1, for example) and not
the external power supply ground when Vee voltage is supplied from J2.
Vee voltage MUST NOT exceed 15 V or the overvoltage protection circuit on
the Vee bus will trip.
3.3.3.5 Regulated DC From On-Board Regulator (Socket U6)

A power supplylvoltage regulator EVM can be installed in platform socket U6
to provide a a regulated Voo voltage (3.3 V or 5 V typical) for audio evaluation
modules installed on the platform that require it. The regulator EVM uses
power from the Vee bus as an input and provides the appropriate regulated
voltage to the platform Voo line.
Voo voltage also appears at screw terminals J6, where it can be used as a
source of regulated power for off-board use, subject to the.maximum current
capabilities of the regulator module installed in U6 and the platform Vee
supply.
Do not allow the Vee voltage to exceed the maximum specified forthe installed
power supplylvoltage regulator EVM.
3.3.3.6 Regulated External DC at Screw Terminals J6

Regulated voltage (3.3 V or 5 V typical - 6 V maximum) from an external
source can be connected to screw terminals J6 to supply the platform Voo line
for audio evaluation modules installed on the platform that require a regulated
Voo supply.
Voltage applied to screw terminals J6 MUST be of the correct polarity and
MUST NOT exceed 6.1 V or the power protection circuit on the Voo line will
trip.

6-30

Details

Inputs and Outputs

3.4 Inputs and Outputs
TI Plug-N-Play Audio Amplifier Evaluation Platform is equipped with several
standard conectors for audio inputs and outputs.

3.4.1

Inputs
In most cases, audio signals enter the platform through either a pair of RCA
phono jacks (J3 and J5) or a miniature (1/8") stereo phone jack (J4). Certain
EVMs, however, may have additional signal input connectors mounted on the
module circuit board.
The platform audio signal input jacks (J3, J4, and J5) are of the closed-circuit
type and are interconnected such that the stereo phone jack is in series with
the RCA phono jacks, and the signal lines are grounded when no plugs are
inserted (Figure 3-5).

Figure 3-5. Platform Audio Input Jacks

Audio
Input

J3DRV~=j~
~L .... ::·m~
J4

~

S i g n a l : S2
Conditioning

...----+

L
L

•

.. ----------,

R"Amplifiers
L

On

'--_ _ _ _ _ _--' Conditioning

J5

The internal switches in the RCA phono jacks (J3 and J5) connect the signal
lines to ground when a plug is not inserted. The internal switches in the stereo
phone jack (J4) connect the module signal inputs to the RCA phono jacks
when a plug is not inserted in the stereo phone jack. These connectors operate
as follows:
•

With no plugs inserted, the signal lines to the inputs of the signal
conditioning socket, U1*, are shorted to ground.

•

With plugs inserted into the RCA phono jacks (J3 and J5) only, the
signal from the phono plugs is routed through the stereo phono jack
internal switches and then on to socket U1*.

•

With a plug inserted into the stereo phone jack (J4), the RCA phono
jacks are disconnected from the input and the signal from the phone
plug is applied to socket U1*.

* or to power amplifier sockets if 82 is set to OFFto bypass conditioning

Details

6-31

Inputs and Outputs

3.4.2 Outputs
Amplified audio signals leave the platform through left and right RCA phono
Jacks (J7 and J9), left and right pairs of compression connectors for stripped
speaker wires (Ja), and a capacitively-coupled miniature (1/a") stereo phone
jack (J10) for headphones (Figure 3-6).

Figure 3-6. Platform Audio Output Jacks and Connectors
J7
r~·---,·-~--·

.'

Amplifier'
or'

ili.~.~. ~7·"···~
J3 J4 J5

Audio
Input

•

'U1

: . 'Slgnll'

."." • .i. ~'_';" _

3.4.2.1

,

. :
j·~Itfo!1lng •
I

iii".

. ····02·.
".
StereoPow«
0 - -.....-

,

:S2
On

"

L'

,

-+----,....

.....

_R_-I-_ _~--=:.j. .

0'4 alidior 03,"-'- - t - -____-~.
Mono 'POwer" .o--L.....+-_._+----'-1.....
AmpIHler(8)

Speaker

Outputs

I

_ _ ·a _ _ • • • • • ,

J9
L

Power Amplifier OutputS/Jacks
The audio output lines from the power amplifiers are separate all the way to
the edge of the platform (output jacks J7, Ja, and J9) - the Out -lines from
the power amplifier sockets are not tied to each other or to platform ground.
This allows certain power amplifier EVMs to operate in various output drive
modes, including some highly-efficient bridged configurations. To reduce
possible emissions, limit the length of speaker wiring to 1 meter or less.

3.4.2.2 Headphone Amplifier Output/Jack
The headphone jack (J1 0) is a stereo miniature phone jack that is capacitively
coupled (via 470 I1F electrolytics) to S3. Source select switch S3 connects the
headphone jack to the output lines of either the headphone amplifier socket
U5, or to the output lines of the power amplifier sockets (U2, U3, and U4).
Some of the TI power amplifier EVMs that can be installed in sockets U2, U3,
or U4 normally operate in the single-ended output mode, and some have the
ability to switch from a bridged output mode to single-ended in response to a
mode control signal. S3 should not be set to the power amplifier position unless
power amplifiers that can operate in the single-ended mode are installed.
When S3 is set to the power amplifier position (U2 - U4), the headphone jack
is connected to the power amplifier Out + output lines. When a headphone plug
is Inserted into the jack, these output lines are returned to the common platform
ground inside J1 0, requiring single-ended power amplifier outputs. For power
amplifier modules that have selectable output modes, a. switch inside the
headphone jack sends a control signal to the power amplifier sockets that can
select the single-ended output mode when a headphone plug is inserted.

6-32

Details

Troubleshooting

3.5 Troubleshooting
This section covers some of the possible difficulties that might be encountered
with platform operation.

o

o

o

o

The platform is connected to an external power source for vee and a
voltage regulator EVM is installed in U6. Neither LED is lit and the EVM
modules are not receiving power.
•

Check that platform power switch S1 is set on ON

•

Check that JP1 or JP2 or JP3 is jumpered and corresponds to the
power source

•

Check fuse F1; replace it if it is found to be open

•

Check platform power switch S1; replace it if it is found to be faulty

The platform is connected to an external power source for Vee and a
voltage regulator EVM is installed in U6. Only LED1 (Vee) is lit. There is
no Voo at JPS and the installed EVMs do not function properly.
•

Check that the voltage regulator EVM is fully seated in socket U6 and
that none of the pins are bent over

•

Substitute a known-good voltage regulator EVM for the module in U6

The platform is connected to an external power source for VOO at J6. LED2
(Voo) is dark and Voo is not reaching the EVMs.
•

Check for correct Voo input supply voltage

•

Check fuse F2; replace it if it is found to be open

Power amplifier EVMs installed in U2, U3, and/or U4 are powered correctly but produce no sound.
•

o

Consult the User's Guide for the installed power amplifier and
determine 1) if the EVM is mute active-high or mute active-low, and 2)
which pin on the module is the mute control input. Measure the voltage
at the mute control input pin of the installed module with no plug
inserted in headphone jack J 10. Ifthe EVM is mute active-high and the
mute pin of the EVM measures Voo, the EVM is being held in the mute
mode. Jumper the other pin on JP8 to reverse the mute control line
polarity.

The power amplifier EVM installed in US is powered correctly, but there is
no sound from headphones when plugged into headphone jack J 10.
•

Check that the headphone jack source select switch (S3) is set to the
U5position

•

Consult the User's Guide for the installed power amplifier and
determine 1) if the EVM is mute active-high or mute active-low, and 2)
which pin on the module is the mute control input. Measure the voltage
at the mute control input pin of the installed module with the
headphone plug inserted in jack J10. If the EVM is mute active-high
and the mute pin of the EVM measures Voo, the EVM is being held in
the mute mode. Jumper the other pin on JP7 to reverse the mute
control line polarity.

Details

6-33

Parts List

3.6 Parts List
Table 4-2. Plug-N-Play Audio Amplifier Evaluation Platform Parts List
Ref

I

DescriPtion·

I

Source

Part No.

Bl

Battery,9·V

Cl

Capacitor, Aluminum, 470 IlF, 25 V

Digi-Key

P5704·ND

C2

Capacitor, Aluminum, 470 IlF, 16 V

Digi-Key

P6230·ND

C3

Capacitor, Aluminum, 470 IlF, 16 V

Digi-Key

P6230-ND

Dl

Diode, Rectifier, 3 A, 50 V

Mouser

583-1N54oo

D2

Diode, Rectifier, 3 A, 50 V

Mouser

583-1N54oo

D3

Diode, Rectifier, 3 A, 50 V

Mouser

583-1N54oo

D4

Diode, Rectifier, 3 A, 50 V

Mouser

583-1N5400

Fl

Fuse, Pico II, 3 A, 125 V, Fast-acting

Littelfuse

251-003

F2

Fuse, Pico II, 3 A, 125 V, Fast-acting

Littelfuse

251-003

Jl

Connector, 2-pin, screw connector, 0.2" centers

Mouser

506-2MV02

J2

Jack, Power, 2.1 mm, PC mount

Mouser

163-5004

J3

Phone Jack, switched, PC mount

Mouser

16PJ396

J4

Phone Jack, Stero, 1/8"

Mouser

161-3504

J5

Phone Jack, switched, PC mount

Mouser

16PJ396

J6

Connector, 2-pin, screw connector, 0.2" centers

Mouser

506-2MV02

J7

Phone Jack, switched, PC mount

Mouser

16PJ396

J8

Connector, 4-pin

Radio Shack

274-622A
16PJ396

J9

Phone Jack, switched, PC mount

Mouser

J10

Phone Jack, 1/8" with SPDT switch

Mouser

161-3503

JPI - JP8

Reader, 2-pin, 100-mil centers, 0.23" top, 0.22" bottom

Digi-Key

SI022-36-ND

LEDI

LED, TI-314, Org, 25-mA

LED2

LED, Tt-314, Red, 25-mA

Rl

Resistor, CF, 430 Ohm, 1/2 W, 5%

R2

Resistor, CF, 150 Ohm, 1/4 W, 5%

R3

Resistor, CF, 240 Ohm, 1/4 W, 5%

R4

Resistor, CF, 1.0 K Ohm, 1/4 W, 5%

R5

Resistor, CF, 1.0 K Ohm, 1/4 W, 5%

SI

Switch, DPDT, 0.2-A, 30-V, pc mount

Digi-Key

EGI908-ND

S2

Switch, DPDT, 0.2-A, 30-V, pc mount

Digi-Key

EGI907-ND

S3

Switch, DPDT, 0.2-A, 30-V, pc mount

Digi-Key

EGI907-ND

VRI

Diode, Zener, 15 V, 1 W, 5%, DO-41

Diodes, Inc.

lN4744A

VR2

Diode, Zener, 6.2 V, 1 W, 5%, DO-41

Diodes, Inc.

lN4735A

XBl

Battery Holder, 9-V, pc mount

Keystone

1294K

Socket Pins, 0.022"-0.032" (Qty: 106) Mil-Max #0295-0-

Digi-Key

ED5008-ND

Standoff, Nylon, 0.375"/6-32 (Qty: 6)

Digi-Key

8441BK-ND

Screw, 0.25"/6-32 (Qty: 6)

Digi-Key

SHUNT, black, closed top (Qty: 3)

Mouser

151-8010

SHUNT, red, open top (Qty: 3)

Mouser

151-8003

PCB

6-34

Printed Circuit Board, 2-layer

SLOP097

Details

Platform EVM Socket Pinouts

3.7 Platform EVM Socket Pinouts
Figure 3-7. Signal Conditioning Socket U1 Pinout
Signal Conditioning
000
<

8

G>
~

<

g

vccO
VDDO
GNDO

....

C

o Righlln

RighlOUIO
GNDO

OGND

LeftOulO
GNDO

OGND

-------****CAUTION****

N/C

N/C
N/C

N/C

~

~

0

g

o Left In

o
o NlC
o
o

~

0

::l

a.

g.
2.
::l

'"

0

Do not insert or remove N/C 0
EVM boards with power
applied
NlC 0
NlC

0

Figure 3-8. Power Amp/fier Socket U2 Pinout

Audio
Power
Amps

r----- -- - - - - - - - -- - - -- - - - - - -._--..,

o
o

Righi In (HP)

000

0
RighlOul- 0
RighlOul+

GNDO

OGND
Righi In (line)

N/C
Mode

OGND

~

ON/C

0
LeftOul- 0

Left In (line)

OGND
Leftln(HP)

N/CO

Mule

OGND

o
o

8

ao

g)
0

0(5

000

GNDO
LeftOul+

0

~-----------------------------~

Details

6-35

Platform EVM Socket Pinouts

Figure 3-9. Power Amplfier Socket U3/U4 Pinout

o
o

<
c
c

Gl

OGND

.---

8

z
c

NlCO
Out-O

NlC

ONIC

o
o

00

0
Mute

.:; ':dNDO
Out +0

Mode

C

In+

C

~
~

OGND

NlCO

Figure 3-10. Headphone Amplfier Socket US Pinout

o
o

Shutdown

Rlghtln

000
Gl

i§

z

c

c

OGND

C'i

0

LeftOut 0

c

GNDO

C

'"

CJ1

o
o
Gl

-l

0

E'i]

Leftln
GND

...

11=-1!2

NlCO
GNDO

NlC

c
'l'
c

~ iij

Left Out 0

~

a

..

Figure 3-11. Power Supply/Regulator Socket U6 Pinout

ONIC

0
<

~

OO"tl

CO

"tI:E

OVCCln

"tim

OVCCln
OGND
OGND

~

~

I

~:IJ

ai

VDDOutO
VDDOutO
GNDO
GNDO

ill

~

~
@

DC

Power
In/Out
+

-I

i

3.35

I

3.3

~

I

~

3.25

3.2
3.15
0

0.5

1
1.5
2
10 - Output Current - A

2.5

3

Figure 1-8. Output Voltage Vs Output Cu"ent (5-V Mode)
OUTPUT VOLTAGE
vs
OUTPUT CURRENT (5-V MODE)

5.15

,

r--.I -.,........-,--...,---.--,

VCC=9V
5.11---+---+---+--+----+---1

>
I

I)

~

'!Do5

8
I

~

5.05

5

4.95

10 - Output Current - A

6-50

Test Results

Figure 1-9. Output Voltage Vs Supply Voltage (3.3-V Mode)
OUTPUT VOLTAGE
VB
SUPPLY VOLTAGE (3.3-V MODE)
3.26

3.255

>
I

t
~

i

8
I

~

10 = 0.25 A

3.25

3.245

--

10=2.5A

- 1-'---

3.24

-~-I"""'

1--

3.235

~-

3.23

5

6

7
8
9
10
11
Vee - Supply Voltage - V

12

13

Figure 1-10. Output Voltage Vs Supply Voltage (5-V Mode)
OUTPUT VOLTAGE
vs
SUPPLY VOLTAGE (5-V MODE)
4.92

-

4.915

>

!

I

J

10 = 0.25 A

4.91

I

~

S 4.905

So

6I

~

4.9

-...-

1-'-

\

4.895

.-----

1--

10=2.5A

4.89
5

6

7

8

9

10

11

12

13

Vee - Supply Voltage - V

Hardware

6--51

Test Results

Figure 1-11. Efficiency Vs Output Current (5-V Mode)
EFFICIENCY
vs
OUTPUT CURRENT
93
92
91

90

I
I,

89

#.

88

~
c

87

1

I

j

85

I

84

-- ---

3.3V

--

I'--..

-...

i
I

82

f

81

80

----

~V

/

86

83

6-52

/

VCC=9V
"I

o

0.5

1.5
10 - Output Current - A

2

2.5

Chapter 2

Design
Procedure
The SLVP097 evaluation module provides a method for evaluating the performance of the TPS2817 MOSFET driver and the TL5001 PWM controller. The
TPS2817 contains all of the circuitry necessary to drive large MOSFETs, including a voltage regulator for higher voltage applications. This section explains how to construct basic power conversion circuits including the design
of the control chip functions and the basic loop. This chapter includes the following topics:

Topic

Page
/,.

2.1 ,'1ntroductlon ;, .• , ........................ ~ ... ;,. ;; .• ; .. ;'.;.;

,'-

~",

•

2;2 'OpentUngSI*lflcatlons ......•........•....•..••••••• :>",,;~, .~
'2;3Def.1ign Pl'ocedure ..... , .. ; ................. ~ .... ~
,:: .•• :'~!:

6-53

Introduction

2.1

Introduction
The SLVP097 is a dc-dc buck converter module that provides a 5-V or 3.3-V
output at up to 2.5 A with an input voltage range of 5.5 V to 12 V. The controller
is a TL5001 PWM operating at a nominal frequency of 275 kHz. The TL5001
is configured for a maximum duty cycle of 100 percent and has short-circuit
protection built in. Output voltage selection is implemented with jumper JP1.

6-54

Operating Specifications

2.2 Operating Specifications
Table 2-1 lists the operating specifications for the SLVP097.

Table 2-1. Operating Specifications
SpeCification

Min

Input Voltage Range

4.5t

Typ

Max

Units

12.6

V

Output Voltage Range
5-V Mode

4.7

5.0

5.3

V

3.3-V Mode

3.1

3.3

3.5

V

2.6

A

Output Current Range

0

Operating Frequency

275

Output Ripple
Efficiency

kHz

50
85%

mV

90%

t For 3.3 V only. minimum input voltage for 5 V output is 5.5 V.

Design Procedure

6-55

Design Procedures

2.3 Design Procedures
Detailed steps in the design of a buck-mode converter may be found in
Designing With the TL5001C PWM Controller (literature number SLVA034)
from Texas Instruments. This section shows the basic steps involved in this
deSign, using the 3.3-V output mode.

2.3.1

Duty Cycle Estimate
The duty cycle for a continuous-mode step-down converter is approximately:

0= Vo +Vd
V I - V SAT
Assuming the commutating diode forward voltage Vd = 0.5 V and the power
switch on voltage VSAT= 0.1 V, the duty cycle for Vj = 5.5,9, and 12 V is 0.70,
0.42, and 0.32, respectively.

2.3.2

Output Filter
A buck converter uses a single-stage LC filter. Choose an inductor to maintain
continuous-mode operation down to 6 percent of the rated output load:
~IO

=2

x 0.06 x

10 = 2 x 0.06 x 2.5 = 0.30 A

The inductor value is:
(VI - V SAT - Vo) x 0 x t

~I

L=

o

(12 - 0.1 - 3.3)

x

=

0.32
0.30

x

(3.63

x

10-6)
= 33.3 ILH

Assuming that all of the inductor ripple current flows through the capacitor and
the effective series resistance (ESR) is zero, the capacitance needed is:
C

~I

=

8 x f x

0

(~VO)

=

0.3

= 2.73

ILF

8 x (275 x 103 ) x 0.05:

Assuming the capacitance is very large, the ESR needed to limit the ripple to
50mVis:
~V

ESR = ~ = 0.05 = 0.167 g
~IO
0.3
The output filter capacitor should be rated at least ten times the calculated
capacitance and 30-50 percent lower than the calculated ESA. This design
used a 220-ILF OS-Con capacitor in parallel with a ceramic to reduce ESA.

2.3.3

Power Switch
Based on the preliminary estimate, rOS(ON) should be less than 0.1 0 V + 2.5 A
= 40 mg. The IRF7406 is a 30-V p-channel MOSFET with rOS(ON) = 40 mg.
Power dissipation (conduction + switching losses) can be estimated as:
Po =

(16 x rDS(ON) x D) + (0.5 x Vi x 10 x tr+f x f)

Design Procedures

Assuming total switching time, tr+f' =100 ns, a 55°C maximum ambient temperature, and rOS(ON) adjustment factor (for high temperature) = 1.6, then:
P D = [2.5 2 x (0.04 x 1.6) x 0.7]

+

[0.5 x 5.5 x 2.5 x (0.1 x 10- 6) x (275 x 103 )]

= 0.41

W

The thermal impedance RaJA = 90°C/W for FR-4 with 2-oz. copper and a oneinch-square pattern, thus:
TJ

2.3.4

= T A + (RaJA

x P D)

= 55 + (90

x 0.41)

= 92°C

Rectifier
The catch rectifier conducts during the time interval when the MOSFET is off.
The 30WQ04 is a 3.3-A, 40-V rectifier in- a D-Pak power surface-mount
package. The power dissipation is:
P D = 10 x V D(1 - DMin ) = 2.5 x 0.6 x 0.68 = 1.02 W

2.3.5 Snubber Network
A snubber network is usually needed to suppress the ringing at the node where
the power switch drain, output inductor, and the rectifier connect. This is
usually a trial-and-error sequence of steps to optimize the network; but as a
starting pOint, select a snubber capaCitor with a value that is 4-1 0 times larger
than the estimated capacitance of the catch rectifier. The 30WQ04 has a
capacitance of 110 pF, resulting in a snubber capaCitor of 1000 pF. Then,
measuring a ringing time constant of 20 ns, R is:
R = 20 x 104:1 =
20 x 104:1 = 20 Q
C
1000 x 10-12
A 22-0 resistor is used in the design.

2.3.6 Controller Functions
The controller functions, oscillator frequency, soft-start, dead-time control,
short-circuit protection, and sense-divider network are discussed in this
section.
The oscillator frequency is set by selecting the resistance value from the graph
in Figure 6 of the TL5001 data sheet. For 275 kHz, a value of 30.1 kQ is
selected.
Dead-time control provides a minimum off-time for the power switch in each
cycle. Set this time by connecting a resistor between DTC and GND. For this
deSign, a maximum duty cycle of 100% is chosen. Then R is calculated as:

+ 1.25 kQ)[ D(VO(1 00%) - VO(o%Y + VO(O%)]
kQ + 1.25 kQ)[1 (1.4 - 0.6) + 0.60] = 44 kQ => 47

R = (ROSC

=

(30.1

Design Procedure

kQ

6-57

Design Procedures
Soft-start is added to reduce power-up transients. This is implemented by
adding a capacitor across the dead-time resistor. In this design, a soft-start
time of 5 ms is used:
C

t

= -1L = 0.005 s = 0 1 J.tF
ROT

47 kg

.

The TL5001 has short circuit protection (SCP) instead of a current sense circuit. If not used, the SCP terminal must be connected to ground to allow the
converter to start up. If a timing capacitor is connected to SCP, it should have
a time constant that is greater than the soft-start time constant. This time
constant is chosen to be 75 ms:
C(J.tF)

2.3.7

= 12.46 x

tscp

= 12.46 x

0.075 s

= 0.93 J.tF

Loop Compensation
Loop compensation is necessary to stabilize the converter over the full range
of load, line, and gain conditions. A buck-mode converter has a two-pole LC
output filter with a 40-dB-per-decade rolloff. The total closed-loop response
needed for stability is a 20-dB-per-decade rolloff with a minimum phase margin
of 30 degrees over the full bandwidth for all conditions. In addition, sufficient
bandwidth must be designed into the circuit to assure that the converter has
good transient response. Both of these requirements are, met by adding
compensation components around the error amplifier to modify the total loop
response.
The first step in design of the loop compensation network is the design of the
output sense divider. This sets the output voltage and the top resistor
determines the relative size of the rest of the compensation design. Since the
TL5001 input bias current is 0.5 J.tA (worst oase) , the divider current should be.
at least 0.5 mAo Using a 1-kg resistor for the bottom of the divider gives a
divider current of 1 mAo Since this is a dual-voltage output, the divider must be
selectable. For a 5-V output, the divider was set for 1 kg and 4 kg. The bottom
of the divider is calculated for the 3-V mode as:
R

=

R
T
Vo - V REF

= 34.3~1

= 1.74 kg

The pulse-width modulator gain can be approximated as the change in output
voltage divided by the change in PWM input voltage:

IN

ApWM

= AVC~MP = 1.~:g.6 = 11.25=21

dB

The LC filter has a double pole at:
1r.-;:;- = 1.87 kHz
2ltv'LC
and rolls off at 40-dB per decade after that until the ESR zero is reached at:
1
=
1
= 26.8 kHz
2ltR ESRC
2lt(0.027)(220 x 10-6)

6-58

Design Procedures

This information is enough to calculate the required compensation values.
Figure 2-1 shows the power stage gain and phase plots.

Figure 2-1. Power Stage Response
FREQUENCY RESPONSE
50

o

40

-45

I

30

I
III

."

20

,

10

I

c

~

-90

1\

I

-20
-30

103

102

10

-180

i

-270

\
\

e.
I

\

-10

-135

I
-225

\

0

t
!

if

-315

/

-360

104

105

Frequency - Hz

Figure 2-2 shows the required error amplifier compensation response.

Figure 2-2. Required Compensation Response
BODE PLOT
90

40

i\
35
30

I
III

15

~

10

\

,

~

I
I

"

,../

103

i

!
I

10

I

-30

!
.c

I

-50

0
10

10

I

5

-5

30

I

20

I

50

"'\

25

."

c

70

~

II.

70
-90

105

Frequency - Hz

This response can be met with the following:
A pole at zero to give high de gain
Two zeroes at 1.87 kHz to cancel the LC poles
A pole at 26.8 kHz to cancel the ESA zero
A final pole to roll off high-frequency gain above 100 kHz

o
o
o
o

Design Procedure

6-59

Design Procedures

The sum of the gains of the modulator, the LC filter, and the error amplifier
needs to be 0 dB at the selected unity-gain frequency of 20 kHz. The modulator
and LC filter gain is -14 dB. The two zeroes at 1.87 kHz in the compensation
network that cancels the LC poles will have a total gain of 41.2 dB at 20 kHz.
Therefore, the pole at zero frequency needs to fumish 0-(-14+41.2) =
-27.2 dB (voltage gain = 0.04365) at 20 kHz. R5 and C12 provide this pole.
R6 is already chosen as 4 kQ. Calculate C12 as:
C12

+ C11

= (2,.;)(f)(R6)(R!qUired Gain)

In practice C12 is much greater than C11, therefore:
C12

= (2,.;)(20 kHZ)(~

kg)(O.04365)

= 0.045

I1F Use C12

= 0.047 I1F

R4 provides the first zero at the LC break point:

1

Use R4

R4 = (2,.;)(1.87 kHz)(C12) = 1.89 kg

1.8 kg

C13 provides the other zero at the LC break point:

1
C13

_

= (1.87 kHz)

1

(20 kHz)
2:n:(R6)

= 0019
.

11

F

Use C13 = 0.018 I1F

R5 provides the compensation for the ESR zero:
R5

= (2:n:)(26.8 ~HZ)(C13) = 330

g

Finally, C11 provides a rolloff filter at high frequency, chosen at 100 kHz:
C11 = (2,.;)(1001kHZ)(R4) = 0.00088 I1F

6-60

Use C11 = 1000 pF

Tone Control Evaluation Module
User's Guide

Uterature Number: SLOU031
January 1999

•
TEXAS
INSTRUMENTS

PrInted on Recycled Paper

6-61

IMPORTANT NonCE

Texas Instruments and its subsidiaries (TI) reserve the right to make changes to their products or to discontinue
any product or service without notice, and advise customers to obtain the latest version of relevant information
to verify, before placing orders, that information being relied on is current and complete. All products are sold
subject to the terms and conditions of sale supplied at the time of order acknowledgement, including those
pertaining to warranty, patent infringement, and limitation of liability.
TI warrants performance of its semiconductor products to the specifications applicable at the time of sale in
accordance with Tl's standard warranty. Testing and other quality control techniques are utilized to the extent
TI deems necessary to support this warranty. Specific testing of all parameters of each device is not necessarily
performed, except those mandated by government requirements.
CERTAIN APPLICATIONS USING SEMICONDUCTOR PRODUCTS MAY INVOLVE POTENTIAL RISKS OF
DEATH, PERSONAL INJURY, OR SEVERE PROPERTY OR ENVIRONMENTAL DAMAGE ("CRITICAL
APPLICATIONS;. TI SEMICONDUCTOR PRODUCTS ARE NOT DESIGNED, AUTHORIZED, OR
WARRANTED TO BE SUITABLE FOR USE IN LIFE-SUPPORT DEVICES OR SYSTEMS OR OTHER
CRITICAL APPLICATIONS. INCLUSION OFTI PRODUCTS IN SUCH APPLICATIONS IS UNDERSTOOD TO
BE FULLY AT THE CUSTOMER'S RISK.
In order to minimize risks associated with the customer's applications, adequate design and operating
safeguards must be provided by the customer to minimize inherent or procedural hazards.
TI assumes no liability for applications assistance or customer product design. TI does not warrant or represent
that any license, either express or implied, is granted under any patent right, copyright, mask work right, or other
intellectual property right of TI covering or relating to any combination, machine, or process in which such
semiconductor products or services might be or are used. TI's publication of information regarding any third
party's products or services does not constitute Tl's approval, warranty or endorsement thereof.

Copyright © 1998, Texas Instruments Incorporated

Preface
Related Documentation From Texas Instruments
•

TI Plug-N-Play Audio Amplifier Evaluation Platform (literature
number SLOU011) provides detailed information on the evaluation
platform and its use with TI audio evaluation modules.

•

TLC2274 Advanced LinCMOS RAIL-TO-RAIL OPERATIONAL
AMPLIFIERS (literature number SLOS190) This is the data sheet
for the TLC2274 Quad operational amplifier integrated circuit used
in the Tone Control EVM.

•

TLV2231 Advanced LinCMOS RAIL-TO-RAIL LOW-POWER
SINGLE OPERATIONAL AMPLIFIER (literature number
SLOS158) This is the data sheet for the TLV2231 operational
amplifier integrated circuit used in the Tone Control EVM.

FCC Warning
This equipment is intended for use in a laboratory test environment only. It generates, uses, and
can radiate radio frequency energy and has not been tested for compliance with the limits of
computing devices pursuant to subpart J of part 15 of FCC rules, which are designed to provide
reasonable protection against radio frequency interference. Operation of this eqUipment in other
environments may cause interference with radio communications, in which case the user at his
own expense will be required to take whatever measures may be required to correct this
interference.

Trademarks
TI is a trademark of Texas Instruments Incorporated.

6-64

Contents
1

Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ..
1.1
Feature Highlights .........................................................
1.2
Description. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ..
1.3 Tone Control EVM Specifications ............................................

6-67
6-68
6-69
6-70

2

Quick Start .....................................................................
2.1
Precautions ...............................................................
2.2
Quick Start List for Platform .................................................
2.3
Quick Start List for Stand-Alone .............................................

6-71
6-72
6-73
6-74

3

Details .........................................................................
3.1
Precautions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ..
3.2
The Tone Control Evaluation Module .........................................
3.2.1 Tone Control EVM Circuit Description .................................
3.2.2 Tone Control EVM Frequency Response ...........................•..
3.3
Using The Tone Control EVM With the Plug-N-Play Evaluation Platform ..........
3.3.1 Installing and Removing EVM Boards .................................
3.3.2 Signal Routing .....................................................
3.3.3 Mute/Mode/Etc.. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ..
3.3.4 Power Requirements. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ..
3.3.5 Inputs and Outputs .................................................
3.4
Using The Tone Control EVM Stand-Alone . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ..
3.5
Tone Control Evaluation Module Parts List ....................................

6-75
6-76
6-77
6-79
6-80
6-81
6-81
6-82
6-83
6-84
6-84
6-85
6-86

6--65

Figures
1-1
2-1
2-2
3-1
3-2
3-3
3-4
3--5

3-6

3-7
3-8

The Tone Control Evaluation Module ...........................................
Quick Start Platform Map .....................................................
Quick Start Module Map - Stand-Alone ........................................
The TI Plug-N-Play Audio Amplifier Evaluation Platform ...........................
Tone Control EVM ............................................................
Tone Control EVM Schematic Diagram ..........................................
Tone Control Evaluation Module Frequency Response ............................
Tone Control EVM Block Diagram ..............................................
Bass and Treble Tone Control Response . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ..
Platform Signal Routing and Outputs ...........................................
Tone Control EVM Stand-Alone Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . ..

6-69
6-72
6-74
6-76
6-77
6-78
6-79
6-79
6-80
6-82
6-85

Tables
3-1

6-66

Tone Control EVM Parts List ................................................... 6-86

Chapter 1

Introduction
This chapter provides an overview of the Texas Instruments (TITM) Tone
Control Evaluation Module (SLOP109). It includes a list of EVM features, a
brief description of the module illustrated with a pictorial diagram, and a list of
EVM specifications.

Topic
1.1

Page
Feature HlghHghts .............................. ;. ~. . . • . • . .• 6-68

n' .• , ..'.......... : ...............

6-69

Tofte Control evMSpeClfIcalIOns ••• " .............. ,~ ..........

6-70

1.2' De$CftptIOJ:! ••• : ••••••

1.3

Ow • • •

6-67

Feature Highlights

1.1

Feature Highlights
The TI Tone Control Evaluation Module and the TI Plug-N-Play Audio Amplifier
Evaluation Platform include the following features:

o

o

o

o

6-68

Tone Control Evaluation Module
•

Individual slide pots for left and right channel volume control

•

Individual slide pots for bass and treble - the bass control adjusts
both channels simultaneously and the treble control adjusts both
channels simultaneously

•

20-d8 cut and 15-d8 boost for both bass and treble

•

3.3-V and 5-V operation

Quick and Easy Configuration with The TI Plug-N-Play Audio Amplifier
Evaluation Platform
•

Evaluation module is designed to simply plug into the platform,
automatically making all signal, control, and power connections

•

Platform provides flexible power options

•

Jumpers on the platform select power and module control options

•

Switches on the platform route signals

•

Platform provides quick and easy audio input and output connections

Platform Power Options
•

External5-V -15-V DC Vee supply inputs

•

External regulated Voo supply input

•

Socket for onboard 5-V/3.3-V VDO voltage regulator EVM

•

Onboard overvoltage and reverse polarity power protection

Platform Audio Inpu1 and Output Connections
•

Left and right RCA phono jack inputs

•

Miniature stereo phone jack inpu1

•

Left and right RCA phono jack outputs

•

Left and right compression speaker terminal outputs

•

Miniature stereo headphone jack output

Introduction

Description

1.2 Description
The Tone Control Evaluation Module is a complete audio volume level and
base and treble control board that is designed primarily for use with the TI
Plug-N-Play Audio Amplifier Evaluation Platform. It consists of separate slide
pots for the right- and left-channel volume control, a slide pot for controlling the
bass response of both channels, a slide pot for adjusting the treble response
of both channels, a single-channel operational amplifier IC, a quad operational
amplifier IC, and a small number of passive components mounted on a circuit
board that measures approximately 21/4 inches by 13/4 inches (Figure 1-1).

Figure 1-1. The Tone Control Evaluation Module
[]l[]

~~~'e IC=====::::{[§JI====:J

C14

Max

R13
[]l[]

~

~s c::=:[§J~======::J

4' c.
'"

C3

~~~~~~~~~~~~~~
oo~~o

~CD;;;"'''''
0+

o

R In

:;;
0+

o
Lin

g

iii

fIl

~ ~ cO

~ m~~

~

:;:

~. ~ iii

.... EI

~
[]l[]

~ Rl

RTVol

I

o

~:

~&l ..~
RB

0
..

LT Vol

~ iii

[]l[]

[]I] []l[] []I]
R5
R6

~
I

~

cO EI

~~

iil ~

'-

+0'"
0

~~.~~

[§J;J:==:::::J
R2

Vdd

G~

~

+0
0
R Out
ij;

+~
LOut

0

Tone Control Board

Single in-line header pins extend from the underside of the module circuit
board to allow the EVM to be plugged into the TI Plug-N-Play Audio Amplifier
Evaluation Platform, orto be wired directly into existing circuits and equipment
when used stand-alone.
The platform, with room for a single tone control evaluation module, is a
convenient vehicle for evaluating Tl's audio power amplifier and related
evaluation modules. The EVMs simply plug into the platform, which
automatically provides power to the modules, interconnects them correctly,
and connects them to a versatile array of standard audio input and output jacks
and connectors. Easy-to-use configuration controls allow the platform and
EVMs to quickly model many possible end-equipment configurations.
There is nothing to build, nothing to solder, and nothing but the speakers
included with the platform to hook up.

Introduction

6-69

Tone Control EVM SpecifICations

1.3 Tone Control EVM Specifications
Supply voltage range, Voo ............................................... 3 V to 5.5 V
Supply current, 100 ..................................................... 6.85 rnA max
Audio input voltage, VI .................................................... 4 Vpp max
Audio output voltage, Vo .................................................. 4 Vpp max

6-70

Introduction

Chapter 2

Quick Start
Follow the steps in this chapter to quickly prepare the Tone Control EVM for
use. Using the Tone Control EVM with the TI Plug-N-Play Audio Amplifier
Evaluation Platform is a quick and easy way to connect power, signal, and
control inputs, and signal outputs to the EVM using standard connectors.
However, the Tone Control EVM can be used stand-alone by making
connections directly to the module pins, and can be wired into existing circuits
or equipment.

Topic

Page

L... ,~. ~~ .. ,15-:72

2.1

P r e c a u t i o n s : ...:';........

2.2

QufCJ($tart Ust forPlatfonn

••~; •••• l'.~,••• ;. ';";";';;;~,~S.:,,;Yf-73

2.3

QuICk Start Ust for Stand-Alone

.................. " ... : •• ~:.. &.-74

6-71

Precautions

2.1

Precautions

Figure 2-1. Quick Start Platform Map

@

f@

DC

Power
In/Out

Power
Input

•

L

i§1n

8

0
0
0
0
0

0000

o
o

Audio
Input

000

L

8

Speaker
Output

0
0
0
0

""'I
;Jl

o

Mode
Mute

+

r--~:;o~o~o;::;--'" ~IP~~rttyl.g>
8
5.!!! ""' HI ""'~
o

c

en

8

00

0

Iii ~ ::!I

;g,g

IIIJ 8~0

0
g>'5!!l
0!ii

z" =----........
°li
Gl-l

6-72

~o

0008

o

o

J

01G

000

III

R3+

____

I\>

HPOut '-

0

R4____ -IIIII__ Rsl

Quick Start

Quick Start List for Platform

2.2 Quick Start List for Platform
Follow these steps when using the Tone Control EVM with the TI Plug·N·Play
Audio Amplifier Evaluation Platform (see the platform user's guide, SLOU011,
for additional details). Numbered callouts for selected steps are shown in
Figure 2-1, and details appear in Chapter 3.

o

Platform preparations

1) Ensure that all external power sources are setto OFFand thatthe platform
power switch 81 is set to OFF.
2) Install the tone control module in the Signal Conditioning platform socket
U1, taking care to align the module pins correctly.
3) Set switch 82 to ONto select signal conditioning by the Tone Control EVM.

4) Install power amplifiers and/or a headphone amplifier module in the
appropriate platform sockets (see the amplifier module User's Guide for
details).
5) Set platform jumpers and switches in accordance with the user's guide for

each amplifier module installed on the platform.

o

Power supply

6) Select and connect the power supply (ensure power supply is setto OFF):

a) Connect an external regulated power supply setto 5 V to platform VOO
power input connector JS taking care to observe marked polarity,
or
b) Install a voltage regulator EVM (SLVP097 or equiv.) in platform socket
US. Connect a 7 V -12 V power source to a platform Vee power input
J1 or J2 and jumper the appropriate power input (see platform user's
guide).

o

Inputs and outputs

7) Ensure that the audio signal source level is set to minimum.
8) Set the EVM right and left volume slide pots to minimum.
9) Connect the audio source to left and right RCA phono jacks J3 and J5 or
stereo miniature phone jack J4.
10) Connect 3-0 - 8-0 speakers to left and right RCA jacks J7 and J9 or to
stripped wire speaker connectors J8, or plug headphones into J10.

o

PowerUp

11) Verify correct voltage and input polarity and set the external power supply
to ON. If Vee and an on board regulator EVM are used to provide Voo, set
platform power switch 81 to ON.
Platform LED2 should light indicating the presence of Voo, and the evaluation
modules installed on the platform should begin operation.
12) Adjust the signal source and Tone Control EVM audio levels as needed.

Quick Start

6-73

Quick Start List for Stand-Alone

2.3 Quick Start List for Stand-Alone
Follow these steps to use the Tone Control EVM stand-alone or when
connecting it into existing circuits or eqUipment. Connections to the tone
control module header pins can be made via individual sockets, wirewrapping, or soldering to the pins, either on the top or the bottom of the module
circuit board. The Tone Control EVM is shown in Figure 2-2 and details appear
in Chapter 3.

Figure 2-2. Quick Start Module Map - Stand-Alone

~r:!le
:~

=====:::[C§JJ:====:::::J
c::::=[§J=======:J

IJII[J

1:1

C14

Max

---+

R13
IJII[J

C3
IJII[J
R31J11[J

1;;

Vdd

+0
0
GND

ij ij .ij~ ~
~ ~

+0

1JII[J:!1 1iil
o

u:
fG
0+
0

Lin

I

0

IJII[J~
I

-Max

RTVol

0

ROut

1;;

+0

o

LOut

t:;::::====::::([§JI==::::J
0

o

R1

CD~

R2

LTVol

Tone Control Board

Power supply

1) Ensure that all external power sources are set to OFF.

2) Connect an external regulated power supply set to 5 V to the module VDD
and GND pins taking care to observe marked polarity.

o

Inputs and outputs

3) Ensure that audio signal source level adjustments are set to minimum.

4) Set the Tone Control EVM volume slide pots to minimum.
5) Connect the audio source to the module R IN and L IN pins, taking care
to observe marked polarity.

o

PowerUp
6) Verify correct voltage and input polarity and set the external power supply
to ON.

The EVM should begin operation.

7) Adjust the signal source and Tone Control EVM audio levels as needed.

6-74

Quick Start

Chapter 3

Details
This chapter provides details on the Tone Control EVM, the steps in the QuickStart List, additional application information, and a parts list for the Tone Control evaluation module.

Topic

Page

·3.1

Precautions ......... ; ................................. ~ • • •• 6-76

3;2

The'Tbne Controt Evaluation Module .;....................... 6-77

3.3, Using the Tone Control EVM WHh the Plug-N-Play
Evaluation Platform •• ; ........................................ 6-81
13.4U$lng The 'Tbne Control EVM Stand~Alone •.•.••••••••••• ~ .•••. &.85

3.5.

Tone Control EValUation Module Parts List .. ; ...........

'c'.,..

6-86

6-75

Precautions

3.1

Precautions

Figure 3-1. The TI Plug-N-Play Audio Amplifier Evaluation Platform

••

o
o
o

Power
Input

.---

L

<,..

8g

000

Rl

Audio 0
Power

L

o
o

f(l~o

00

0

0

0

8

~
~

noo
l"8

8

0

:=====:::::::~
~
000
0
00

J- ~ 8

8

o

8

"'P-IU-9.-N-.p-la-Y-A-U-di-o-Am-PI-ifi-er-..J

6-76

~

8

8

Speaker
Output

0
0
0
8

0
0008
C
ol-c..IMode +
""'0
dl Mute
o
_~o:;:::;:::;::;----'::t: PolarHy C/l
r0 0 0
~ILOljc
0
c.!!!
HI
0
'" c..
c...'l'
0
c
51 ~
;B~

o

····CAUTION····
0
Do not insert or remove 0
EVM boards with power
applied
0

Evaluation Platform
SLOP097 Rev. C.1

0
000 8 1---0
0

§ 8

8

~-----

o
o

000

Amps 0
r-'-- 0

o
o
o

Power
In/Out

-g
c..
o
."
o
en

Signal Conditioning
000

Audio
Input

DC

8

tn

§ ~~J

Gl-l
z " 1,.;0'--_ _ _---' ~/6

0';

:s

8~

R3+
_
__

'" HPOut 0

R4_ _ _ ---RS

Details

The Tone Control Evaluation Module

3.2 The Tone Control Evaluation Module
The Tone Control Evaluation Module provides a convenient way to control the
audio volume and the tonal response of audio amplifier EVMs plugged into the
TI Plug-N-Play Audio Amplifier Evaluation Platform. Tone controls allow the
frequency response of the audio system to be adjusted to compensate for the
response of speakers and their enclosures, or to simply provide a more
pleasing sound. A pair of slide pots adjusts the volume of each channel
independently, while a single slide pot adjusts the bass response of both
channels simultaneously and another slide pot adjusts the treble response of
both channels. The module provides a gain of 2 at the maximum volume
setting when both tone controls are at their midpoints (flat).
Although the Tone Control EVM is designed to be used with the TI Plug-N-Play
Audio Amplifier Evaluation Platform (Figure 3-1), it can be wired directly into
circuits or equipment. The module has single in-line header connector pins
mounted to the underside of the board. These pins allow the module to be
plugged into the TI platform, which automatically makes all the signal input and
output, power, and control connections to the module.
The module connection pins are on O.1-inch centers to allow easy use with
standard perf board and plug board-based prototyping systems. Or, the EVM
can be wired directly into existing circuits and equipment when used
stand-alone.
The module appears in Figure 3-2 and its schematic is shown in Figure 3-3.

Figure 3-2. Tone Control EVM

~~~Ie CI=====:I[§J~==:::J
Max

--

:~ c:::::::[§J~======::J

III
C14
R13

III

=:
0+
~
0
Rln

i

c..

'"
0+
0

Lin

Details

s-n

The Tone Control Evaluation Module

Figure 3-3. Tone Control EVM Schematic Diagram
VDD
VDD

C5
0.1 I1F

R3

20kn

mid

R4
20kn

T

U1 =nC2274
Quad Op-Amp
U2=nV2231
Single OpoAmp

-::-

GND
-::R19
10kn

mid

R10A
100kn

C7
0.033J1F
R14
10kn

R16
3.3kn

C15
1.0l1F

tr·~

R19
100kn

R12

mid

Lin

R10B
100kO

R15
10kn
R17
2.2kn
C16
1.0 J1F

tr

R20
100kn
mid

LOut

The Tone Control EVM is a variation of the classic and very popular Baxandall
negative feedback tone control. This circuit allows a range of adjustment from
cut, through flat, to boost In bass response with a single potentiometer.
Another potentiometer provides the same range of adjustment for the treble
response. The component values indicated in the schematic provide the
response curve shown in Figure 3-4. Each of the tone adjusting
potentiometers is a dual unit, allowing the simultaneous adjustment of both
channels with a single control. A separate volume control for each channel
allows the adjustment of balance between the channels as well as volume.
A single TLC2274 quad rail-to-rail operational amplifier IC contains all the
amplifiers required for both channels. A TLV2231 operational amplifier IC is
connected to provide a midpoint voltage (and signal ground) for proper
operation of the TLC2274.

6-78

Details

The Tone Control Evaluation Module

Figure 3-4. Tone Control Evaluation Module Frequency Response
20

15
III

-a

5

!I

0

-

Full Boost

I

10

I

i

I

roo... ~
,~

1'0""

'5

!-5

~

~

~

,

./

-15
-20 i-oo'

Flat

"'

/

-10

~

~ ....

~

,

-

Full Cut

15

1K

100

10K

20K

f - FrequenCy - Hz

3.2.1

Tone Control EVM Circuit Description
Each of the two separate channels on the Tone Control EVM is basically an
active filter built around an IC operational amplifier. An active filter design was
chosen over a passive filter circuit because active filters have the
frequency-response adjusting components located in the feedback loop of the
filter amplifiers, providing much lower THO, little or no insertion loss, and a
symmetrical response about the axis in both boost and cut, compared with
most passive designs. Each channel also includes an input buffer amplifier to
provide some gain, isolation from source impedance variations, signal
inversion, and a low-impedance drive for the filter circuit.
A block diagram of the right channel of the Tone Control EVM is shown in
Figure 3-5. The left channel is identical.

Figure 3-5. Tone Control EVM Block Diagram
C1
U2:A

In

Feedback Network

R1 ~_--I
(Log)

Mid

Buffer AmplHler

U2:D

>--+--+1 Out
Tone Control

Filter AmplHIer

The input buffer amplifier provides a gain of approximately 2 (RF/R'N) with the
resistor values installed on the module. Input capacitor C1 blocks DC and sets
the overall low-frequency rolloff of the EVM at approximately 16 Hz with the
installed value of 2.2 IlF. Volume control R1 has an audio taper to provide a
perceived response in volume that is proportional to the physical position of
the slider and gives an adjustment range at the output of the buffer amplifier
of from 0 V to approximately 2X the audio signal input voltage.

Details

6-79

The Tone Control Evaluation Module

The tone adjusting action in each channel of the Tone Control EVM is provided
by an equalized amplifier (or active filter) created by placing a frequencydependent negative feedback network around an operational amplifier.
Almost any overall gain-versus-frequency characteristic can be defined by the
design of the feedback network.
The EVM provides the familiar Hi-Fi tone control, in which the low audio
frequencies can be boosted or cut approximately 20 dB with the bass control
and the high audio frequencies can be boosted or cut approximately 20 dB with
the treble control. Middle frequencies are not affected by the tone controls. An
overall flat response (no boost or cut at frequency extremes) is obtained when
the tone controls are at their mid-point position.

3.2.2 Tone Control EVM Frequency Response
The overall Tone Control EVM frequency response can be shifted up or down
by changing the values of capacitors C7, Cg, C11, and C12 in the tone
adjusting networks on the module. Care must be taken, however, because the
surface-mount solder pads on the board are somewhat fragile and will not
survive a large number of soldering/desoldering operations.
To shift the EVM frequency response downward, for example, increase the
values of the capacitors in the tone adjusting networks. Doubling the values
of C7, Cg, C11, and C12 shifts the break frequency downward a full octave
(Case B, Figure H). Conversly, halving the values of C7, Cg, C11, and C12
shifts the break frequency upward a full octave. .
Note that to keep the boost and cut break frequencies the same, the value of
C7 must equal that of Cg, and the value of C11 must equal that of C12. In
addition, although the bass and treble break frequencies can be adjusted
separately if desired, to maintain the overall shape and symmetry of the
response, all four capacitors must be increased or decreased by the same
factor.

Figure

~.

Bass and Treble Tone Control Response
20

......

C~~: C~, 6e ~ ~~ri~ I1F
Cll, C12 =3300 pF

,,~

15
III

'i'

5

j

0 ,..-

v

Gi

Ca~B

r-..

'5
~-5

-10

-20

. ~~ .....
.... t""""
. .. :.r;:..

:;....-

....- i.-'

ioOi'

CaseA -

..~~

v

{

. ~~

......

ce

Case B: C7,
= O.068I1F
C11, C12= 6800 pF
I

15

----

~

..:"'0

!"'"

~:/

o

-15

.... .. ~""
. ~ r-..

1"'::0-

10

100

.1

I

I

LIlli

1K

10K

20K

f - Frequency - Hz

Details

Using The Tone Control EVM With the Plug-N-Play Evaluation Platform

3.3

Using The Tone Control EVM With the Plug-N-Play Evaluation Platform
The Tone Control Evaluation Module was designed to be used with the TI
Plug-N-Play Audio Amplifier Evaluation Platform. It simply plugs into socket
U1.
The following paragraphs provide additional details for using the Tone Control
EVM with the platform.

3.3.1

Installing and Removing EVM Boards
TI Plug-N-Play evaluation modules use single-in-line header pins installed on
the underside of the module circuit board to plug into sockets on the platform.
The EVM pins and the platform sockets are keyed such that only the correct
type of EVM can be installed in a particular socket, and then only with the
proper orientation.
Evaluation modules are easily removed from the platform by simply prying
them up and lifting them out of their sockets. Care must be taken, however,
to prevent bending the pins.

3.3.1.1

EVM Insertion
1) Remove all power from the evaluation platform.
2) Locate the appropriate socket on the platform.
3) Orient the module correctly.
4) Carefully align the pins of the module with the socket pin receptacles.
5) Gently press the module into place.
6) Check to be sure that all pins are seated properly and that none are bent
over.

3.3.1.2 EVM Removal
1) Remove all power from the evaluation platform.
2)

Using an appropriate tool as a lever, gently pry up one side of the module
a small amount.

3) Change to the opposite side of the module and use the tool to pry that side
up a small amount.
4) Alternate between sides, prying the module up a little more each time to
avoid bending the pins, until it comes loose from the socket.
5)

Lift the EVM off of the platform.

Details

6-81

Using The Tone Control EVM With the Plug-N-Play Evaluation Platform

3.3.2

Signal Routing
Signal flow on the platform is controlled by two signal routing switches, as
shown in Figure 3-7.

Figure 3-7. Platform Signal Routing and Outputs
...-_ _ _ _ _ _... Off

R

~

~

r·:-·~--:~~::7/·'·~·~·_

....._____~

......-....--i' .. '
. ,~! R
R
,. . >,·,u~> •.:,:...--+-----~
. ;AmPi~":~"", ';-.'_ - ! -_ _ _ _-+

'-0:'-..-,.- ". ,..

Audio
Input

<~i . ·'· ..

On

::"

L

+

J7,J8,J9
Speaker
Outputs

L

. ""\;,.;......-+-----~ +
• .,;" •.••,.i .• ·;'" .,,;,;'.".

U2-U4

J10
Headphone
Output

3.3.2.1

Signal Conditioning
The Tone Control EVM plugs into the Signal Conditioning socket (U1) on the
platform. The audio signal from the platform input jacks can be applied to the
signal conditioning socket (U1) or can bypass socket U1 as determined by
conditioning switch S2.

o

Switch S2 selects the tone control signal conditioning or bypasses it

3.3.2.2 Headphone Output Jack
Switch S3 is the source selectfor the stereo headphone output jack, J 10. The
headphone jack is capacitively coupled (via 470 J.LF electrolytics) and can
output either the signal from the headphone amplifier in socket U5, or the
signal from the power amplifier installed in sockets U2 - U4, as determined
by the setting of headphone source select switch S3.
When S3 is set to the power amplifier position (U2 - U4), the headphone jack
is connected to the power amplifier OUT+ output lines. When a plug is inserted
into the jack, signals output through J10 are returned to platform ground,
requiring single-ended power amplifier operation. A switch inside the
headphone jack produces a control signal that can be routed to the power
amplifier socket to shut down the power amplifier EVM or switch it to
single-ended output mode when a plug is inserted.
See the User's Guide for the power amplifier and/or the headphone amplifier
installed on the platform for information on the correct setting of switch S3.

6-82

Details

Using The Tone Control EVM With the Plug-N-Play Evaluation Platform

3.3.3

Mute/Mode/Etc.
Some power amplifier EVMs have a mute or mode control input pin. This
allows the power amplifier to enter the mute state for decreased power
consumption or to switch output modes in response to a control signal applied
to this pin.
In typical applications, as often found in notebook computers, portable audio
products, and such, the internal speakers mute when headphones are
plugged into the headphone jack, or internal speakers mute when external
speakers are connected. In applications using separate speaker and
headphone amplifiers, the power amplifier can be shut down (muted) to
conserve power when the headphone amplifier is in use.
Output mode switching allows some power amplifier EVMs to operate in the
bridge-tied load (BTL) output mode for increased power to internal speakers
and then switch to single-ended mode to drive headphones when a plug is
inserted into the headphone jack, eliminating the need for a separate
headphone amplifier.
The platform is equipped with mute/mode control signal select and polarity
jumpers and a headphone source switch to provide the maximum flexibility in
configuring the operation of the various power amplifier and headphone
amplifier EVMs that might be installed on the platform. See the User's Guide
for the power amplifier and/or the headphone amplifier installed on the
platform for information on the correct settings of platform mute, mode,
polarity jumpers, and the platform headphone source switch.

Details

6-83

Power Requirements

3.3.4

Power Requirements
The Tone Control Evaluation Module can operate from any voltage between
approximately 3 V and 5.5 V. For best performance (highest output power with
lowest distortion), the module should be operated at approximately 5 V unless
there is a specific reason for operating it from a lower voltage.
The TI Plug-N-Play Audio Amplifier Evaluation Platform with a voltage
regulator EVM installed on it can provide a regulated Voo supply from a wide
variety of unregulated Vee voltage inputs between approximately 5.5 V and
12 V, including an on board 9 -V battery. Or, an external regulated power source
can be used to supply Voo voltage to the platform and the tone control
evaluation module installed on it.
Although the Tone Control EVM draws a very small amount of current from the
supply, power amplifiers installed on the platform can draw as much as
approximately 2 A from the power supply during continuous full power output.
Any power supply connected to the platform should be capable of providing
adequate current to the power amplifier installed on the platform to avoid
clipping of the output Signal during peaks. Current consumption driving
speakers at normal listening levels is typically 0.5 A or less.
The platform is equipped with overvoltage and reverse-polarity supply voltage
input protection in the form of fused crowbar circuits.

3.3.5

o

Voo voltage applied to platform screw terminals J6 MUST NOT exceed
the absolute maximum rating for any EVM installed on the platform, or
damage may result. In no case should Voo voltage ofthe incorrect polarity
or in excess of 6.1 V be applied to screw terminals J6 of the platform, or
the power protection circuit on the Voo line will trip.

o

Vee voltage applied to the platform MUST NOT exceed the maximum
voltage input specified for the voltage regulator module installed in socket
U6 (12 V for the SLVP097), or damage to the voltage regulator module
may result. In no case should Vee voltage applied to the platform exceed
15 V, or the overvoltage protection circuit on the Vee bus will trip.

Inputs and Outputs
The TI Plug-N-Play Audio Amplifier Evaluation Platform is equipped with
several standard conectors for audio inputs and outputs.

3.3.5.1

Inputs
Audio signals enter the platform through either a pair of RCA phono Jacks (J3
and J5) or a miniature (1/8") stereo phone jack (J4). The platform audio signal
input jacks (J3, J4, and J5) are of the closed-circuit type, grounding the signal
input lines when no plugs are inserted.

3.3.5.2 Outputs
Amplified audio output signals leave the platform through left and right RCA
phono jacks (J7 and J9), left and right pairs of compression connectors for
stripped speaker wires (J8), and optionally, through a miniature (1/8") stereo
phone jack (J1 0), for headphones.

6-84

Details

Using The Tone Control EVM Stand-Alone

3.4 Using The Tone Control EVM Stand-Alone
Using the Tone Control Evaluation Module stand-alone is much the same as
using it with the platform. The same 5-V power supply requirement exists.

3.4.1

Tone Control EVM Connected for Stand-Alone Operation

Figure 3-8. Tone Control EVM Stand-Alone Operation

=====::([§Jl======::J

Il!IIIJ

~s c=::=@J~======::J

Il!IIIJ

~':Ie

--

C::I

C14

Max

R13

C3

Il!IIIJ
R31l!111J

~-t--+--+-<:J: +
Input ~
(Right)
Audio

Rln

Audio
Ii;;
Input :>-+-+-;-8 +
(Left)

Lin

-=-

~

IiiiI
EI

00:11:110

:11

c;;

:;;:

co '"

0> ;;;

fIl

•

i!il~~~

I

0

1l!IIIJ2

~.:11

IiiiI

" " ' " El

R8

"'ElIl!lllJ Il!IIIJ Il!IIIJ

[§J

Rl

RTVoI

R2

LTVol

I

0

iG iiiiiii IiiiI &l

R5

RS

[§J

Max

~

Vdd!;;

+

G--+---<

GND

~~.~bl ~
i! !i?

Il!IIIJ
o
en

Audio
Output
(Right)

+ G-+--+-t---3~

~~

~~
~~

ROut

&i
+ e--+--+-+-~

Audio
Output
(Left)

Tone Control Board

Details

6-85

Tone Control Evaluation Module Parts Ust

3.5 Tone Control Evaluation Module Parts List
Table 3-1. Tone Control EVM Parts List
Reference

Description

Size

EVM
3

Sourcel
Part Number

Qty.

Cl, C2, C4

Capacitor, ceramic, 2.2J!F, 16 V, YV5

1206

C3

Capacitor, ceramic, 10 J!F, 16 V, YV5

1210

C15,C16

Capacitor, ceramic, 1 J!F, 16 V, YV5

1206

2

TDK
C3216Y5V1Cl05Z

C7, C8, C9, Cl0

Capacitor, ceramic 0.033 J!F, 50 V, NPO

1206

2

Digi-Key

C5,C6

Capacitor, ceramic, 0.1 J!F, 50 V, X7R

1206

2

Digi-Key
PCC104BCT-ND

Cll, C12, C13,
C14

Capacitor, ceramic 3300 pF, 50 V, NPO

1206

4

Digi-Key

Rl0,R18

Dual potentiometer, 100 kn, linear taper,
slide control

2

CTS 448XC351109

Rl, R2

Potentiometer, 50 kn, audio taper, slide
control

2

CTS 448XC3503BAN

R5, R6, R9, Rll,
R12, R13, R14,
R15

Resistor, CF, 10 kn, 1/8 W, 5%

1206

8

R3,R4,R7,R8

Resistor, CF, 20 kn, 1/8 W, 5%

1206

4

R16,R17

Resistor, CF, 3.3 kn, 1/8 W, 5%

1206

2

R19,R20

Resistor, CF, 10 kn, 1/8 W, 5%

1206

2

Jl-J5

Header, 2 position, 100-mil centers

Ul

TLV22311DBV IC operational amplifier

U2

TLC2274CD quad IC operational amplifier

PCB

PCB, Tone Control EVM

6-86

TDK
C3216Y5V1C225Z
TDK
C3216Y5V1Cl06Z

5

Digi-Key S1022-36ND

SOT-23

TI

SOIC

TI
TISLOP109

Details

7-1

Contents
Page
Mechcanical Data ......................................................... 7-3

s::

(I)

n
:r
Q)

_.

~

nQ)

-c

a
Q)

7-2

MECHANICAL DATA

o (R-POSO-G··)

PLASTIC SMALL-QUTLINE PACKAGE

14 PIN SHOWN

0 1r.050 (1,27)

14

0.020(0,51)
0.014 (0,35)

8

I'IT
~I
I
0.010 (0,25) ®

-------r-r
--r

0.244 (6,20)
0.228 (5,80)

I

0.157 (4,00)
0.150 (3,81)

l---------.l~

LA

7

rfiULiUiidiid~

t

0.069 (1,75) MAX

0.010 (0,2;J
0.004 (0,10)

~

8

14

16

A MAX

0.197
(5,00)

0.344
(8,75)

0.394
(10,00)

A MIN

0.189
(4,80)

0.337
(8,55)

0.386
(9,80)

DIM

4040047/010/96
NOTES: A.
B.
C.
D.

All linear dimensions are in inches (millimeters).
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusion, not to exceed 0.006 (0,15).
Falls within JEDEC MS-012

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALLAS, TEXAS 75265

7-3

MECHANICAL DATA

MECHANICAL INFORMATION
PowerPADTM PLASTIC SMALL-oUTLINE PACKAGE

DCA (R·PDSo-G**)
48 PINS SHOWN

ThennalPad
(See Note D)

,---------------1
I
I
I

I
~

7,90

I
IL _______________ I
~

o

1~._1______________

2~4

A ______________

~boooooooooooooooooooooood~
~
~
0,05

-r-;,20 MAX

~

-~0~_hL
1~10,10 ~

48

56

64

A MAX

12,60

14,10

17,10

A MIN

12,40

13,90

16,90

DIM

4073259/A 01198
NOTES: A.
B.
C.
D.

All linear dimensions are in millimeters.
This drewing is subject to change without notice.
Body dimensions do not include mold flash or protrusion not to exceed 0,15.
The package thermal performance may be enhanced by bonding the thermal pad to an extemal thermal plane.
This pad is electrically and thermally connected to the backside of the die and possibly selected leads.
E. Falls within JEDEC MQ-I53

PowerPAD is a trademark of Texas Instruments Incorporated.

7-4

-!111ExAs
INSTRUMENTS
POST OFFICE BOX 855303 • OALLAS, TEXAS 75265

MECHANICAL DATA

MECHANICAL INFORMATION

DGN (8-PDSo-G8)

PowerPADTM PLASTIC SMALL·OUTLINE PACKAGE

r- 1 r trsl~1 0,25@1
Thermal Pad

(See Note D)

,----,
I
I
I

I

I

I

3,05
2,95

,-O.,.....,...L"T'"-"T'"-..,..-..,..---.J..,....,..J

J

4,98
4,78

C~_4~--------L
2,95

__ '

iE!i1ij E1 a~
(d+--)_~
eMAX
~J 1c>IO'10~
4073271/A04198

NOTES: A.
B.
C.
D.

All linear dimensions are In millimeters.
This drawing is subject to change without notice.
Body dimensions Include mold flash or protrusions.
The package thermal performance may be enhanced by attaching an extemal heat sink to the thermal pad.
This pad is electrically and thermally connected to the backside of the die and possibly selected leads. The dimension of the thermal
pad is 68 mils (height as illustrated) x 70 mils (width as illustrated) (maximum). The pad is centered on the bottom of the package.
E. Falls within JEDEC MQ-187

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DALI.AS, TEXAS 75265

7-5

MECHANICAL DATA

MECHANICAL INFORMATION
DGQ (9-PDSo-G10)

PowerPADTM PLASTIC SMALL-oUTLINE PACKAGE

Thermal Pad
(See Note D)

,----,

I
I
I

I

I

I

4,98
4,78

3,05
2,95

...O..,..,....,L-r-..,.-..,..-,...--.J!""'l""T-'

J

~ .~~5

--!--------L

2,95

__~(J~)_~
1c>IO'10~

4073273/A 04198
NOTES: A.
B.
C.
D.

All linear dimensions are in millimeters.
This drawing Is subject to change without notice.
Body dimensions do not include mold flash or protrusion.
The package thermal performance may be enhanced by bonding the thermal pad to an external thermal plane.
This pad Is electrically and thermally connected to the backside of the die and possibly selectad leads. The dimension of the thermal
pad is 68 mils (height as Illustrated) x 70 mils (width as Illustrated) (maximum). The pad Is centered on the bottom of the package.

PowerPAD Is a trademark of Texas Instruments Incorporated.

~TEXAS

7-6

INSTRUMENTS
POST OFFICE BOX 66530a • DAUAS. TEXAS 75285

MECHANICAL DATA

MECHANICAL INFORMATION
PowerPADTM PLASTIC SMALL·OUTLINE PACKAGE

DWP (R·PDSO·G**)
20 PINS SHOWN

1r-:.

11

1-$-1

0.020 (0,51)
0.010 (0 25)
0.014 (0,35)··
'

®1
.

-------rThermelPad
(SaeNote D)

1-- - - I

o

0.419 (10,65)

I
I
I
I
I
L ____ -.J

0.400 (10,16)
0.299 (7,59)
0.293 (7,45)

I

r---------I~
10

l.bDDDDDDDDDL
0.104 (2,65) MAX

0.006 (0,15)
0.002 (0,05)

seating Plene

J

~.
DIM

16

20

24

28

A MAX

0.410
(10,41)

0.510
(12,95)

0.610
(15,49)

0.710
(18,03)

A MIN

0.400
(10,16)

0.500
(12,70)

0.600
(15,24)

0.700
(17,78)
4147575/A 04198

NOTES: A.
B.
C.
D.

All linear dimensions are in inches (millimeters).
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusion not to exceed 0.006 (0,15).
The package thermal performance may be enhanced by bonding the thermal pad to an extemal thermal plane.
This pad is electrically and thermally connected to the backside of the die and possibly selected leads.

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAllAS, TEXAS 75265

7-7

MECHANICAL DATA

MECHANICAL INFORMATION
PLASTIC DUAL-IN-LiNE PACKAGE

NE (R-PDIP-T**)
20 PIN SHOWN

0.070 (1,78) MAX
11

20

~
MIN

16

DIM

A

B
10

A-----+11 f_~')Y"
J-r-,,,-,.,.-rr-r-r-TT..,.,.....,..'--"-.,.,.....-I

---

0.914 (23,22)

MAX

0.780 (19,80)

0.975 (24,77)

MIN
MAX

---

0.930 (23,62)

---

1.000 (25,40)

0.240 (6,10)

0.260 (6,61)

0.260 (6,60)

0.280 (7,11)

MIN
MAX

C

20

0.200 (5,08) MAX
seating Plane

0.155 (3,94)
0.125 (3,17)

1.-1 0.100 (2,54) 1

14

~f

II

-::-=:::--=1

~ j..- ~:~~! (~:~J 1~ 10.010 (0,25) ® 1

.1C:51)

8

~10.2oo(5~08)MAX
----".----_-"~'-f

~
-..!

1.-1 0.100 (2,54) 1

-J ~

1+-_ _ _-.1-- 0.310 (7,87)
0.290 (7,37)

MIN

Seating Plane

0.155J3,94)
0.125 (3,17)

0.021 (0,533)
0.015 (0,381)

~

1 10:10 (0,25) ® 1
L.I...L-_-'--'---'--"LJ

0.010 (0,25) NOM

JL
4040054/804/95

NOTES: A. All linear dimensions are in inches (millimeters).
B. This drawing is subject to change without notice.
C. Falls within JEDEC M8-001 (16 pin only)

~TEXAS

INSTRUMENTS
7-8

POST OFFICE BOX 655303 • DALLAS. TEXAS 75265

MECHANICAL DATA

MECHANICAL DATA
PowerPADTM PLASTIC SMALL-OUTLINE

PWP (R-PDSo-G**)
20 PINS SHOWN

11

0,30
0,19
11

1-$-1

®I

-----.,..Thermal Pad

,---,
I
I

0,10

L-!..-'--=---~"-'

(See Note D)

4,50
4,30

I

6,60
6,20

""0rT"'TT"T'ILI""TT'"-n--TT-"'-.lM'l""TT"'T'I~ ~

~A

10

r6 0 0 0 0 0 0 0 0 0 3-.J:
~,20 MAX

Q.!§
0,05

J

~

seaUng Plane

1=-1 0,10

r:i'\,--+-_~

_ ~

.

~

14

16

20

24

28

A MAX

5,10

5,10

6,60

7,90

9,80

A MIN

4,90

4,90

6,40

7,70

9,60

DIM

4073225IF 10198
NOTES: A.
B.
C.
D.

All linear dimensions are in millimeters.
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusions.
The package thermal performance may be enhanced by bonding the thermal pad to an external thermal plane.
This pad is electrically and thermally connected to the backside of the die and possibly selected leads.
E. Falls within JEDEC MQ-I53

PowerPAD is a trademark of Texas Instruments Incorporated.

~TEXAS

INSTRUMENTS
POST OFFICE BOX 655303 • DAUAS, TEXAS 75265

7-9

7-10

NOTES

TI Worldwide Technical Support
Internet
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www.ti.com/sc

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+81-3-3344-5317
0120-81-0036
www.ti.com/sc/jpic
www.tij;co.jp/pic

@2oooTexas Instruments Incorporated
Printed in the USA

~1ExAs

INSTRUMENTS

A120799

"'!1

TEXAS
INSTRUMENTS
Printed in U.S .A.
03/00

SLOD004



---

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