Radio Handbook 16 1962
User Manual: Radio-Handbook-16-1962
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This book is revised and brought up
to date (at irregular intervals) os
necessitated by technical progress.
111E
11111110
II
1111ß001i
Sixteenth Edition
WILLIAM
I. ORR, W6SAI
Editor, 16th Edition
The Standard of the Field
for
-
advanced amateurs
practical radiomen
practical engineers
practical technicians
Published and distributed to the electronics trade by
EDITORS and ENGINEERS, Ltd.
Dealers: Electronic distributors, order from us. Bookstores,
Taylor, Hillside, N.J. Export (exc. Canada). order from N.M.
Summerland
www.americanradiohistory.com
,
California
newsdealers order from Baker li
Snyder Co., 440 Park Ave. So., N.Y. 16.
libraria.
THE
HANDBOOK
RADIO
SIXTEENTH
EDITION
Copyright, 1962, by
Editors and Engineers, Ltd.
Summerland, California, U.S.A.
Copyright under Pan -American Convention
All Translation Rights Reserved
Printed in U.S.A.
The "Radio Handbook" is also available on special order in Spanish and
Italian editions; French, German, and Flemish -Dutch editions are in
preparation or planned.
Outside North America, if more convenient, write: (Spanish) Marcombo, S.A., Av.
Jose Antonio, 584, Barcelona, Spain; (Italian) Edizione C.E.L.I., Via Gandino 1,
Bologna, Italy; (French, German, Flemish- Dutch) P. H. Brans, Ltd., 28 Prins Leopold
St., Borgerhout, Antwerp, Belgium.
Other Outstanding Books from the Same Publisher
(See Announcements at Back of Book)
THE RADIOTELEPHONE LICENSE MANUAL
THE SURPLUS RADIO CONVERSION MANUALS
THE SURPLUS HANDBOOK
THE WORLD'S RADIO TUBES
(
RADIO TUBE VADE MECUM)
TILE WORLD'S EQUIVALENT TUBES ( EQUIVALENT TUBE VADE MECUM)
THE WORLD'S TELEVISION TUBES (TELEVISION TUBE VADE MECUM)
www.americanradiohistory.com
THE RADIO
HANDBOOK
16th Edition
Table of Contents
Chapter One. INTRODUCTION TO RADIO
Amateur Radio
-1
Station and Operator Li
-2
The Amateur Bands
-3
Starting Your Study
-4
11
Chapter Two. DIRECT CURRENT CIRCUITS
The Atom
2 -1
Fundamental Electrical Units and Relationships
2 -2
Capacitors
Electrostatics
2 -3
Magnetism and Electromagnetism
2 -4
RC and RL Transients
2 -5
21
Chapter Three. ALTERNATING CURRENT CIRCUITS
Alternating Current
3 -1
41
41
1
1
1
1
-
3 -2
3 -3
3 -4
3 -5
11
12
12
14
21
22
30
35
38
Resonant Circuits
Nonsinusoidal Waves and Transients
53
Transformers
Electric Filters
61
63
Chapter Four. VACUUM TUBE PRINCIPLES
Thermionic Emission
4 -1
4 -2
The Diode
4 -3
The Triode
4 -4
Tetrode or Screen Grid Tubes
Mixer and Converter Tubes
4 -5
Electron Tubes at Very High Frequencies
4 -6
4 -7
Special Microwave Electron Tubes
The Cathode -Ray Tube
4 -8
4 -9
Gas Tubes
4 -10
Miscellaneous Tube Types
58
67
67
71
72
77
79
80
81
84
87
88
Chapter Five. TRANSISTORS AND SEMI -CONDUCTORS
Atomic Structure of Germanium and Silicon
5 -1
Mechanism of Conduction
5 -2
The Transistor
5 -3
Transistor Characteristics
5 -4
Transistor Circuitry
5 -5
Transistor Circuits
5 -6
3
90
90
90
92
94
96
103
Chapter Six. VACUUM TUBE AMPLIFIERS
6 -1
Vacuum Tube Parameters
6 -2
Classes and Types of Vacuum -Tube Amplifiers
6 -3
6 -8
Biasing Methods
Distortion in Amplifiers
Resistance- Capacitance Coupled Audio- Frequency Amplifiers
Video -Frequency Amplifiers
Other Interstage Coupling Methods
Phase Inverters
6 -9
D -C
6 -10
Single -ended Triode Amplifiers
Single -ended Pentode Amplifiers
Push -Pull Audio Amplifiers
Class B Audio Frequency Power Amplifiers
Cathode- Follower Power Amplifiers
Feedback Amplifiers
Vacuum -Tube Voltmeters
6 -4
6 -5
6 -6
6 -7
6 -11
6 -12
6 -13
6 -14
6 -15
6 -16
Amplifiers
106
106
107
108
109
109
113
113
115
117
118
120
121
123
127
129
130
Chapter Seven. HIGH FIDELITY TECHNIQUES
7 -1
The Nature of Sound
7 -2
The Phonograph
7 -3
The High Fidelity Amplifier
7 -4
Amplifier Construction
7 -5
The "Baby Hi Fi"
7 -6
A Transformerless 25 Watt Music Amplifier
134
134
136
138
142
143
146
Chapter Eight. RADIO FREQUENCY VACUUM TUBE AMPLIFIERS
Tuned RF Vacuum Tube Amplifiers
8 -1
Grid Circuit Considerations
8 -2
Plate- Circuit Considerations
Radio- Frequency Power Amplifiers
8 -3
Class C. R -F Power Amplifiers
8 -4
Class B Radio Frequency Power Amplifiers
8 -5
Special R -F Power Amplifier Circuits
8 -6
Class ABI Radio Frequency Power Amplifiers
151
151
151
153
154
154
159
162
166
Chapter Nine. THE OSCILLOSCOPE
9 -1
A Typical Cathode -Ray Oscilloscope
Display of Waveforms
9 -2
9 -3
Lissajous Figures
9 -4
Monitoring Transmitter Performance with the Oscilloscope
9 -5
Receiver I -F Alignment with an Oscilloscope
9 -6
Single Sideband Applications
170
Chapter Ten. SPECIAL VACUUM TUBE CIRCUITS
10 -1
Limiting Circuits
10 -2
Clamping Circuits
10 -3
Multivibrators
10 -4
The Blocking Oscillator
10 -5
Counting Circuits
10 -6
Resistance - Capacity Oscillators
10 -7
Feedback
185
185
187
188
190
190
4
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170
175
176
179
180
182
191
192
194
Chapter Eleven. ELECTRONIC COMPUTERS
Digital Computers
11 -1
Binary Notation
11 -2
Analog Computers
11 -3
-4
-5
11 -6
11 -7
11
11
195
195
197
199
The Operational Amplifier
Solving Analog Problems
Non -linear Functions
Digital Circuitry
200
202
204
Chapter Twelve. RADIO RECEIVER FUNDAMENTALS
Detection or Demodulation
12 -1
Superregenerative Receivers
12 -2
Superheterodyne Receivers
12 -3
Mixer Noise and Images
12 -4
211
Stages
12 -5
R -F
12 -6
Signal- Frequency Tuned Circuits
I -F Tuned Circuits
Detector, Audio, and Control Circuits
12 -7
12 -8
12 -9
12 -10
12 -11
12 -12
Noise Suppression
Special Considerations in
Receiver Adjustment
Receiving Accessories
U -H -F
Receiver Design
Chapter Thirteen. GENERATION OF RADIO FREQUENCY ENERGY
Self -Controlled Oscillators
13 -1
Quartz Crystal Oscillators
13 -2
Crystal Oscillator Circuits
13 -3
Radio Frequency Amplifiers
13 -4
Neutralization of R.F. Amplifiers
13 -5
13 -6
13 -7
13 -8
13 -9
13 -10
13 -11
13 -12
13 -13
13 -14
13 -15
Neutralizing Procedure
Grounded Grid Amplifiers
Frequency Multipliers
Tank Circuit Capacitances
L and Pi Matching Networks
Grid Bias
Protective Circuits for Tetrode Transmitting Tubes
Interstage Coupling
Radio- Frequency Chokes
Parallel and Push -Pull Tube Circuits
Chapter Fourteen.
14 -1
14 -2
14 -3
R -F
FEEDBACK
Feedback Circuits
Feedback and Neutralization of a Two -Stage R -F Amplifier
Neutralization Procedure in Feedback -Type Amplifiers
R-F
Chapter Fifteen. AMPLITUDE MODULATION
.. ..
Sidebands ..
15 -1
Mechanics of Modulation
15 -2
Systems of Amplitude Modulation
15 -3
15 -4
15 -5
205
205
207
208
210
Input Modulation Systems
Cathode Modulation
5
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214
216
223
225
229
233
234
237
237
242
245
249
250
253
256
256
259
263
265
267
268
270
271
272
272
275
277
280
280
281
283
290
295
15 -6
-7
15 -8
15
The Doherty and the Terman-
Woodyard Modulated Amplifiers 296
298
The Bias -Shift Heising Modulator
305
Speech Clipping
Chapter Sixteen. FREQUENCY MODULATION AND RADIOTELETYPE
TRANSMISSION
16 -1
Frequency Modulation
16 -2
Direct FM Circuits
16 -3
Phase Modulation
16 -4
Reception of FM Signals
16 -5
Radio Teletype
Chapter Seventeen. SIDEBAND TRANSMISSION
17 -1
Commercial Applications of SSB
17 -2
Derivation of Single -Sideband Signals
17 -3
Carrier Elimination Circuits
17 -4
Generation of Single -Sideband Signals
17 -5
Single Sideband Frequency Conversion Systems
17 -6
Distortion Products Due to Nonlinearity of R-F Amplifiers
17 -7
Sideband Exciters
17 -8
Reception of Single Sideband Signals
17 -9
Double Sideband Transmission
17 -10 The Beam Deflection Modulator
Chapter Eighteen. TRANSMITTER DESIGN
18 -1
Resistors
18 -2
Capacitors
Wire and Inductors
Grounds
Holes, Leads and Shafts
Parasitic Resonances
Parasitic Oscillation in R-F Amplifiers
Elimination of V -H -F Parasitic Oscillations
Checking for Parasitic Oscillations
18 -3
18 -4
18 -5
18 -6
18 -7
18 -8
18 -9
Chapter Nineteen. TELEVISION AND BROADCAST INTERFERENCE
19 -1
Types of Television Interference
19 -2
Harmonic Radiation
19 -3
19 -4
19 -5
Low-Pass Filters
Broadcast Interference
HI -FI Interference
Chapter Twenty. TRANSMITTER KEYING AND CONTROL
20-1
Power Systems
20 -2
Transmitter Control Methods
20 -3
Safety Precautions
20 -4
Transmitter Keying
20 -5
Cathode Keying
20 -6
Grid Circuit Keying
20 -7
Screen Grid Keying
20 -8
Differential Keying Circuits
6
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308
308
311
315
317
322
323
323
324
328
330
336
340
342
347
349
350
352
352
354
356
358
358
360
361
362
364
367
367
369
372
375
382
383
383
387
389
391
393
394
395
396
Chapter Twenty -One. RADIATION, PROPAGATION AND TRANSMISSION
LINES
21 -1
21
-2
21
-3
399
.._
399
Radiation from an Antenna
General Characteristics of Antennas
.-
400
Radiation Resistance and Feed -Point Impedance
Antenna Directivity
403
409
21 -8
Bandwidth
Propagation of Radio Waves
Ground -Wave Communication
Ionospheric Propagation -_
21 -9
Transmission Lines
416
21 -10
Non -Resonant Transmission Lines
417
21 -11
Tuned or Resonant Lines
420
Line Discontinuities
421
21 -4
21
-5
21
-6
21
-7
21
-12
406
409
410
412
422
Chapter Twenty -Two. ANTENNAS AND ANTENNA MATCHING
End -Fed Half -Wave Horizontal Antennas
22 -1
Center -Fed Half -Wave Horizontal Antennas
22 -2
422
423
__
22 -3
The Half -Wave Vertical Antenna
426
22 -4
The Ground Plane Antenna
427
22 -5
The Marconi
22 -6
Space- Conserving Antennas
430
22 -7
Multi -Band Antennas
Matching Non -Resonant Lines to the Antenna
Antenna Construction
Coupling to the Antenna System
432
Antenna Couplers
A Single -Wire Antenna Tuner
450
22 -8
22 -9
22 -10
22 -11
22 -12
428
Antenna
438
444
447
452
455
Chapter Twenty-Three. HIGH FREQUENCY ANTENNA ARRAYS
Directive Antennas
23 -1
Long Wire Radiators
23 -2
455
457
458
460
23 -3
The V Antenna
23 -4
The Rhombic Antenna
23 -5
Stacked -Dipole Arrays
461
23 -6
Broadside Arrays
464
23 -7
End -Fire Directivity
469
23 -8
Combination End -Fire and Broadside Arrays
471
473
473
Chapter Twenty -Four. V -H -F AND U -H -F ANTENNAS
Antenna Requirements
24 -1
24 -2
Simple Horizontally- Polarized Antennas
475
24 -3
Simple Vertical -Polarized Antennas
24 -4
The Discone Antenna
24 -5
Helical Beam Antennas
476
477
479
24 -6
The Corner -Reflector and Horn -Type Antennas
481
24 -7
VHF
Horizontal Rhombic Antenna
Multi- Element V-H -F Beam Antennas
482
24 -8
_.....
7
www.americanradiohistory.com
484
Chapter Twenty-Five. ROTARY BEAMS
25 -1
Unidirectional Parasitic End -Fire Arrays (Yogi Type)
25 -2
The Two Element
25 -3
The Three -Element
25 -4
Feed Systems
Beam .___
490
490
490
492
_
Array
494
25 -6
for Parasitic (Yogi) Arrays
Unidirectional Driven Arrays
Bi- Directional Rotatable Arrays
25 -7
Construction of Rotatable Arrays
502
25 -8
Tuning the Array
505
25 -9
Antenna Rotation Systems
Indication of Direction
"Three- Band" Beams
509
25 -5
25 -10
25 -11
26 -3
26 -4
26 -5
501
510
510
Chapter Twenty -Six. MOBILE EQUIPMENT DESIGN AND
26 -1
Mobile Reception
26 -2
500
INSTALLATION
Mobile Transmitters
Antennas for Mobile Work
Construction and Installation of Mobile Equipment
Vehicular Noise Suppression
Chapter Twenty-Seven. RECEIVERS AND TRANSCEIVERS
27 -1
Circuitry and Components
27 -2
A Simple Transistorized Portable B -C Receiver
27 -3
27 -4
An Inexpensive Bandpass-Filter Receiver
A Compact Transceiver for 10 and 15 Meters
27 -6
"Siamese" Converter for Six and Two Meters
A Deluxe Mobile Transceiver
27 -7
A Deluxe Receiver for the DX Operator
27 -5
Chapter Twenty- Eight. LOW
POWER
TRANSMITTERS AND EXCITERS
511
511
517
518
520
523
526
529
529
530
539
547
555
564
.... 577
A Transistorized 50 Mc. Transmitter and Power Supply
A Deluxe 200 -Watt Tabletop Transmitter
578
Strip -Line Amplifiers for VHF Circuits
A "9T0" Electronic Key
595
597
Chapter Twenty -Nine. HIGH FREQUENCY POWER AMPLIFIERS
602
602
28 -1
28 -2
28 -3
28 -4
29 -1
Power Amplifier Design
29 -2
Push -Pull Triode
29 -3
Push -Pull Tetrode
29 -4
29 -5
29 -6
29 -7
29 -8
29 -9
29 -10
29 -11
29 -12
29 -13
Amplifiers
Amplifiers
Tetrode Pi- Network Amplifiers
Grounded -Grid Amplifier Design
A 350 Watt P.E.P. Grounded -Grid Amplifier
The "Tri-Bander" Linear Amplifier for 20 -15 -10
An 813 Grounded -Grid Linear Amplifier
The KW -2. An Economy Grounded -Grid Linear Amplifier
A Pi- Network Amplifier for
C -W, A -M, or SSB
Kilowatt Amplifier for Linear or Class C Operation
A 2- Kilowatt P.E.P. All -Band Amplifier
A 3 -1000Z Linear Amplifier
8
www.americanradiohistory.com
581
604
606
609
612
617
622
627
634
643
649
654
661
Chapter Thirty.
SPEECH
AND AMPLITUDE MODULATION EQUIPMENT
669
30 -1
Modulation
669
30 -2
Design of Speech Amplifiers and Modulators
672
Modulator
673
30 -3
General Purpose Triode Class
30 -4
A 10 -Watt Amplifier- Driver
30 -5
A 15 -Watt Clipper- Amplifier
677
678
30 -6
A 200 -Watt 811 -A De -Luxe Modulator
679
30 -7
Zero Bias Tetrode Modulators
683
B
684
Chapter Thirty -One. POWER SUPPLIES
Power Supply Requirements
31 -1
684
-2
Rectification Circuits
689
-3
Standard Power Supply Circuits
690
31 -4
Selenium and Silicon Rectifiers
695
31 -5
100 Watt Mobile Power Supply
31 -6
31 -7
Transistorized Power Supplies
Two Transistorized Mobile Supplies
697
703
706
31 -8
Power Supply Components
31 -9
Special Power Supplies
31 -10
Power Supply Design
31 -11
-_
300 Volt, 50 Ma. Power Supply
1500 Volt, 425 Milliampere Power Supply
716
A Dual Voltage Transmitter Supply
718
718
31
31
31
-12
31
-13
-14
31
707
709
713
.
.
_
A Kilowatt Power Supply
717
720
Chapter Thirty -Two. WORKSHOP PRACTICE
720
723
32 -1
Tools
32 -2
The
32 -3
TVI -Proof Enclosures
724
32 -4
Enclosure Openings
32 -5
Summation of the Problem
725
725
32 -6
Construction Practice
32 -7
Shop Layout
Material
726
729
731
Chapter Thirty- Three. ELECTRONIC TEST EQUIPMENT
Voltage, Current and Power
33 -1
Measurement of Circuit Constants _
33 -2
731
737
33 -3
Measurements with a Bridge
738
33 -4
Frequency Measurements
739
33 -5
Antenna and Transmission Line Measurements
740
33 -6
A Simple Coaxial
742
33 -7
Measurements on Balanced Transmission Lines
33 -8
A
33-9
The Antennascope
33 -10
A Silicon Crystal Noise Generator
747
749
33 -11
A Monitor Scope for AM and SSB
750
"Balanced"
Reflectometer
Chapter Thirty -Four. RADIO
MATHEMATICS AND
744
745
SWR Bridge
CALCULATIONS
9
www.americanradiohistory.com
752
FOREWORD TO THE SIXTEENTH EDITION
Over two decades ago the historic first edition of the RADIO HANDBOOK
was published as a unique, independent, communications manual written
especially for the advanced radio amateur and electronic engineer. Since that early
issue, great pains have been taken to keep each succeeding edition of the RADIO
HANDBOOK abreast of the rapidly expanding field of electronics.
So quickly has the electron invaded our everyday affairs that it is now no
longer possible to segregate one particular branch of electronics and define it as
radio communications; rather, the transfer of intelligence by electrical means
encompasses more than the vacuum tube, the antenna, and the tuning capacitor.
Included in this new, advanced Sixteenth Edition of the RADIO HANDBOOK
are fresh chapters covering electronic computers, r.f. feedback amplifiers, and high
fidelity techniques, plus greatly expanded chapters dealing with semi- conductors
and special vacuum tube circuits. The other chapters of this Handbook have been
thoroughly revised and brought up to date, touching briefly on those aspects in
the industrial and military electronic fields that are of immediate interest to the
electronic engineer and the radio amateur. The construction chapters have been
completely re- edited. All new equipments described therein are of modern
design, free of TV! producing problems and various unwanted parasitic
oscillations.
The writing and preparation of this Handbook would have been impossible
without the lavish help that was tended the editor by fellow amateurs and sympathetic electronic organizations. Their friendly assistance and helpful suggestions
were freely given in the true amateur spirit to help make the 16th edition of the
RADIO HANDBOOK an outstanding success.
The editor and publisher wish to thank these individuals and companies whose
unselfish support made the compilation and publication of this book an interesting and inspired task.
-WILLIAM I. ORR, W6SAI, 3A2AF, Editor
Thomas Consalvi, W3EOZ,
Barker & Williamson, Inc.
Claude E. Doner, W3FAL,
Radio Corporation of
America
John A. Evans, W9HRH,
Potter & Brumfield Co.
Wayne Green, W2NSD,
73 Magazine
Jo Jennings, W6EI,
Jennings Radio Mfg. Co.
E. A. Neal, W4ITC,
General Electric Co.
Harold Vance, K2FF,
Radio Corporation of
America
Blackhawk Engineering Co.
H. E. Blaksley, K7ASK
Byron Hunter, W6VML
Clifford Johnson, WOURQ
Herbert Johnson, W7GRA
Thomas Lamb, K8ERV
James G. Lee, W6VAT
Hugh MacDonald, W6CDT
Otto Miller, K6ENX
Robert Moore, W7JNC
B. A. Ontiveros, W6FFF
(drafting)
A. L. Patrick, W9EHW
Raymond Rinaudo, W6KEV
Robert Sutherland, W6UOV
W. H. Sayer, Jr., WA6BAN
Mel Whiteman, W6BZ
www.americanradiohistory.com
CHAPTER ONE
Introduction to Radio
to the teaching of the principles of equipment
design and signal propagation. It is in response
to requests from schools and agencies of the
Department of Defense, in addition to persistent requests from the amateur radio fraternity,
that coverage of these principles has been expanded.
The field of radio is a division of the much
larger field of electronics. Radio itself is such
a broad study that it is still further broken
down into a number of smaller fields of which
only shortwave or high- frequency radio is covered in this book. Specifically the field of communication on frequencies from 1.8 to 450 megacycles is taken as the subject matter for this
work.
1
The largest group of persons interested in
the subject of high-frequency communication is
the more than 350,000 radio amateurs located
in nearly all countries of the world. Strictly
speaking, a radio amateur is anyone interested
in radio non -commercially, but the term is ordinarily applied only to those hobbyists possessing transmitting equipment and a license from
the government.
It was for the radio amateur, and particularly for the serious and more advanced amateur, that most of the equipment described in
this book was developed. However, in each
equipment group, simple items also are shown
for the student or beginner. The design principles behind the equipment for high- frequency
radio communication are of course the same
whether the equipment is to be used for commercial, military, or amateur purposes, the
principal differences
lying
-1
Amateur Radio
Amateur radio is a fascinating hobby with
many phases. So strong is the fascination offered by this hobby that many executives, engineers, and military and commercial operators
enjoy amateur radio as an avocation even
though they are also engaged in the radio field
commercially. It captures and holds the interest of many people in all walks of life, and in
all countries of the world where amateur activities are permitted by law.
Amateurs have rendered much public service through furnishing communications to and
from the outside world in cases where disaster
has isolated an area by severing all wire com-
munications. Amateurs have a proud record of
heroism and service in such occasion. Many
expeditions to remote places have been kept
in touch with home by communication with amateur stations on the high frequencies. The amateur's fine record of performance with the
"wireless" equipment of World War I has been
surpassed by his outstanding service in World
in construction
practices, and in the tolerances and safety
factors placed upon components.
With the increasing complexity of high-frequency communication, resulting primarily from
increased utilization of the available spectrum, it becomes necessary to delve more deeply into the basic principles underlying radio
War II.
By the time peace came in the Pacific in
the summer of 1945, many thousand amateur
operators were serving in the allied armed
forces. They had supplied the army, navy,
marines, coast guard, merchant marine, civil
service, war plants, and civilian defense organizations with trained personnel for radio,
communication, both from the standpoint of
equipment design and operation and from the
standpoint of signal propagation. Hence, it will
be found that this edition of the RADIO HANDBOOK has been devoted in greater proportion
11
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Introduction to Radio
12
radar, wire, and visual communications and
for teaching. Even now, at the time of this
writing, amateurs are being called back into
the expanded defense forces, are returning to
defense plants where their skills are critically
needed, and are being organized into communication units as an adjunct to civil defense
groups.
1
Station and Operator Licenses
-2
Every radio transmitting station in the
United States no matter how low its power
must have a license from the federal government before being operated; some classes of
stations must have a permit from the government even before being constructed. And every
operator of a transmitting station must have
an operator's license before operating a transmitter. There are no exceptions. Similar laws
apply in practically every major country.
There are at present six
classes of amateur operator licenses which have
been authorized by the Federal Communications Commission. These classes differ in
"Classes of Amateur
Operator Li
many
respects, so each will
be
discussed
briefly.
(a) Amateur Extra Class. This class of license is available to any U. S. citizen who at
any time has held for a period of two years or
more a valid amateur license, issued by the
FCC, excluding licenses of the Novice and
Technician Classes. The examination for the
license includes a code test at 20 words per
minute, the usual tests covering basic amateur
practice and general amateur regulations, and
an additional test on advanced amateur practice. All amateur privileges are accorded the
holders of this operator's license.
(b) General Class. This class of amateur
license is equivalent to the old Amateur Class
B license, and accords to the holders all amateur privileges except those which may be set
aside for holders of the Amateur Extra Class
license. This class of amateur operator's license is available to any U. S. citizen. The
examination for the license includes a code
test at 13 words per minute, and the usual examinations covering basic amateur practice
and general amateur regulations.
(c) Conditional Class. This class of amateur license and the privileges accorded by it
are equivalent to the General Class license.
However, the license can be issued only to
those whose residence is more than 125 miles
airline from the nearest location at which FCC
examinations are held at intervals of not more
than three months for the General Class amateur operator license, or to those who for any
THE
RADIO
of several specified reasons are unable to appear for examination.
(d) Technician Class. This is a new class
of license which is available to any citizen of
the United States. The examination is the same
as that for the General Class license, except
that the code test is at a speed of 5 words per
minute. The holder of a Technician class license is accorded all authorized amateur privileges in the amateur frequency bands above
220 megacycles, and in the 50-Mc. band.
(e) Novice (.lass. this is a new class of
license which is available to any U. S. citizen
who has not previously held an amateur license of any class issued by any agency of
the U. S. government, military or civilian. The
examination consists of a code test at a speed
of 5 words per minute, plus an examination on
the rules and regulations essential to beginner's operation, including sufficient elementary radio theory for the understanding of those
rules. The Novice Class of license affords
severely restricted privileges, is valid for only
a period of one year (as contrasted to all other
classes of amateur licenses which run for a
term of five years), and is not renewable.
All Novice and Technician class examinations are given by volunteer examiners, as regular examinations for these two classes are
not given in FCC offices. Amateur radio clubs
in the larger cities have established examin
ing committees to assist would -be amateurs
of the area in obtaining their Novice and Technician licenses.
1
-3
The Amateur Bands
Certain small segments of the radio frequen-
cy spectrum between 1500 kc. and 10,000 .fc.
are reserved for operation of amateur radio
stations. These segments are in general agreement throughout the world, although certain
parts of different amateur bands may be used
for other purposes in various geographic regions. In particular, the 40 -meter amateur band
is used legally (and illegally) for short wave
broadcasting by many countries in Europe,
Africa and Asia. Parts of the 80 -meter band
are used for short distance marine work in Europe, and for broadcasting in South America.
The amateur bands available to American radio amateurs aree
The 160 -meter band is divided into 25- kilocycle
segments on a regional
basis, with day and night power limitations,
and is available for amateur use provided no
interference is caused to the Loran (Long
Range Navigation) stations operating in this
band. This band is least affected by the 11-
160 Meters
(1800 Kc. -2000 Kc.)
www.americanradiohistory.com
Amateur Bands
HANDBOOK
year solar sunspot cycle. The Maximum Usable Frequency (MUF) even during the years
of decreased sunspot activity does not usually
drop below 4 Mc., therefore this band is not
subject to the violent fluctuations found on
the higher frequency bands. DX contacts on
on this band are limited by the ionospheric
absorption of radio signals, which is quite
high. During winter nighttime hours the absorption is often of a low enough value to permit trans -oceanic contacts on this band. On
rare occasions, contacts up to 10,000 miles
have been made. As a usual rule, however,
160 -meter amateur operation is confined to
ground -wave contacts or single -skip contacts
of 1000 miles or less. Popular before World
War II, the 160 -meter band is now only sparsely occupied since many areas of the country
are blanketed by the megawatt pulses of the
Loran chains.
The 80 -meter band is the
most popular amateur
band in the continental
United States for local "rag- chewing" and
traffic nets. During the years of minimum sunspot activity the ionospheric absorption on
this band may be quite low, and long distance
DX contacts are possible during the winter
night hours. Daytime operation, in general, is
limited to contacts of 500 miles or less. During the summer months, local static and high
ionospheric absorption limit long distance contacts on this band. As the sunspot cycle advances and the MUF rises, increased ionospheric absorption will tend to degrade the
long distance possibilities of this band. At
the peak of the sunspot cycle, the 80 -meter
band becomes useful only for short-haul communication.
80 Meters
(3500 Kc. -4000 Kc.)
The 40 -meter band is high
Kc) enough in frequency to be
severely affected by the
11 -year sunspot cycle. During years of minimum solar activity, the MUF may drop below
7 Mc., and the band will become very erratic,
with signals dropping completely out during
the night hours. Ionospheric absorption of signals is not as large a problem on this band as
it is on 80 and 160 meters. As the MUF gradually rises, the skip- distance will increase on
40 meters, especially during the winter months.
At the peak of the solar cycle, the daylight
skip distance on 40 meters will be quite long,
and stations within a distance of 500 miles or
so of each other will not be able to hold communication. DX operation on the 40 -meter band
is considerably hampered by broadcasting stations, propaganda stations, and jamming trans40 Meters
(7000 Kc. -7300
13
mitters. In Europe and Asia the band is in a
chaotic state, and amateur operation in this region is severely hampered.
At the present time,
20 Meters
(14,000 Kc.-14,350 Kc.) the 20 -meter band is
by far the most popular
band for long distance contacts. High enough
in frequency to be almost obliterated at the
bottom of the solar cycle, the band nevertheless provides good DX contacts during years
of minimal sunspot activity. At the present
time, the band is open to almost all parts of
the world at some time during the year. During the summer months, the band is active until the late evening hours, but during the winter months the band is only good for a few
hours during daylight. Extreme DX contacts
are usually erratic, but the 20 -meter band is
the only band available for DX operation the
year around during the bottom of the DX cycle.
As the sunspot count increases and the MUF
rises, the 20 -meter band will become open for
longer hours during the winter. The maximum
skip distance increases, and DX contacts are
possible over paths other than the Great Circle
route. Signals can be heard the "long paths,"
180 degrees opposite to the Great Circle path.
During daylight hours, absorption may become
apparent on the 20 -meter band, and all signals
except very short skip may disappear. On the
other hand, the band will be open for worldwide DX contacts all night long. The 20 -meter
band is very susceptible to "fade- outs"
caused by solar disturbances, and all except
local signals may completely disappear for
periods of a few hours to a day or so.
This is a relatively
new band for radio
amateurs since it has
only been available for amateur operation
since 1952. Not too much is known about the
characteristics of this band, since it has not
been occupied for a full cycle of solar activity. However, it is reasonable to assume that
it will have characteristics similar to both the
20 and 10 -meter amateur bands. It should have
a longer skip distance than 20 meters for a
given time, and sporadic -E (short -skip) should
be apparent during the winter months. During
a period of low sunspot activity, the MUF will
rarely rise as high as 15 meters, so this band
will be "dead" for a large part of the year.
During the next few years, 15 -meter activity
should pick up rapidly, and the band should
support extremely long DX contacts. Activity
on the 15 -meter band is limited in some areas,
15 Meters
(21,000 Kc.- 21,450 Kc.)
www.americanradiohistory.com
14
I
n t r o d u c
t
i
o n
t o
R a d
i
o
since the older model TV receivers have a
21 Mc. i -f channel, which falls directly in the
15 -meter band. The interference problems
brought about by such an unwise choice of
intermediate frequency often restrict operation
on this band by amateur stations unfortunate
enough to be situated near such an obsolete
receiver.
Meters
(28.000 Kc.- 29,700 Kc.)
10-
During the peak of the
sunspot cycle, the 10meter band is without
doubt the most popular
amateur band. The combination of long skip
and low ionospheric absorption make reliable
DX contacts with low powered equipment possible. The great width of the band (1700 kc.)
provides room for a large number of amateurs.
The long skip(1500 miles or so) prevents nearby amateurs from hearing each other, thus
dropping the interference level. During the winter months, sporadic -E (short skip) signals
up to 1200 miles or so will be heard. The 10meter band is poorest in the summer months,
even during a sunspot maximum. Extremely
long daylight skip is common on this band, and
and in years of high MUF the 10 -meter band
will support intercontinental DX contacts during daylight hours.
The second harmonic of stations operating
in the 10 -meter band falls directly into television channel 2, and the higher harmonics of
10 -meter transmitters fall into the higher TV
channels. This harmonic problem seriously
curtailed amateur 10 -meter operation during
the late 40's. However, with the new circuit
techniques and TVI precautionary measures
stressed in this Handbook, 10 -meter operation
should cause little or no interference to nearby television receivers of modern design.
At the peak of the sunspot
cycle, the MUF occasionally rises high enough to permit DX contacts up to 10,000 miles or so on
6 meters. Activity on this band during such a
period is often quite high. Interest in this band
wanes during a period of lesser solar activity,
as contacts, as a rule, are restricted to short skip work. The proximity of the 6-meter band
to television channel 2 often causes interference problems to amateurs located in areas
where channel 2 is active. As the sunspot cycle increases, activity on the 6 -meter band will
increase.
Six Meters
(50 Mc. -54 Mc.)
The V -HF Bands
(Two Meters and "Up ")
v -h -f bands are
the least affected by
The
the vagaries of the
sunspot cycle and the Heaviside layer. Their
predominant use is for reliable communication
over distances of 150 miles or less. These
T H E
R A D
I
O
bands are sparsely occupied in the rural sections of the United States, but are quite heavily congested in the urban areas of high popu-
lation.
In recent years it has been found that v -h -f
signals are propagated by other means than by
line -of-sight transmission. "Scatter signals,"
Aurora reflection, and air -mass boundary bending are responsible for v -h -f communication up
to 1200 miles or so. Weather conditions will
often affect long distance communication on
the 2 -meter band, and all the v -h -f bands are
particularly sensitive to this condition.
The other v -h -f bands have had insufficient
occupancy to provide a clear picture of their
characteristics. In general, they behave much
as does the 2 -meter band, with the weather
effects becoming more pronounced on the higher frequency bands.
1
-4
Starting Your Study
When you start to prepare yourself for
amateur examination you will find that the
cuit diagrams, tube characteristic curves,
formulas appear confusing and difficult of
derstanding. But after
the
cirand
un-
a few study sessions
becomes sufficiently familiar with the
notation of the diagrams and the basic concepts of theory and operation so that the acquisition of further knowledge becomes easier
and even fascinating.
As it takes a considerable time to become
proficient in sending and receiving code, it is
a good idea to intersperse technical study sessions with periods of code practice. Many
short code practice sessions benefit one more
than a small number of longer sessions. Alternating between one study and the other keeps
the student from getting "stale" since each
type of study serves as a sort of respite from
the other.
When you have practiced the code long
enough you will be able to follow the gist of
the slower sending stations. Many stations
send very slowly when working other stations
at great distances. Stations repeat their calls
many times when calling other stations before
contact is established, and one need not have
achieved much code proficiency to make out
their calls and thus determine their location.
one
The Code
The applicant for any class of amateur operator license must be able
to send and receive the Continental Code
(sometimes called the International Morse
Code). The speed required for the sending and
receiving test may be either 5, 13, or 20 words
per minute, depending upon the class of license, assuming an average of five characters
to the word in each case. The sending and re-
www.americanradiohistory.com
Learning the Code
HANDBOOK
A
6
C
D
E
.
=
MI
N
O
P
Q
MO
H
I
,J
K
L
M
MED
IEM
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7
NEI
EMI
V
W
=El 41
X
MIMEE
Y
MIMED MI
Z
MD
ME Ma
S
U
-.
)
2
3
4
Mo =El
5
6
R
F
G
MD
8
9
GM
.
=..
fm. gm,
OM 4=1
1M
ME, MED 4=1
IMP
0
MEANS ZERO. AND IS WRITTEN IN THIS
WAY TO DISTINGUISH IT FROM THE LETTER 'O''
IT OFTEN IS TRANSMITTED INSTEAD AS ONE
LONG DASH (EQUIVALENT TO 5 DOTS)
0
MI
PERIOD (.)
WAIT SIGN (AS)
COMMA (,)
DOUBLE DASH (BREAK)
INTERROGATION (7)
QUOTATION MARK (")
ERROR (ERASE SIGN)
COLON
(
FRACTION BAR( /)
END OF MESSAGE (AR)
)
SEMICOLON
END OF TRANSMISSION (SK)
INTERNAT. DISTRESS SIG. (SOS)
(I)
PARENTHESIS
15
( I
Figure
.
_
MMD
mo
IMID
e
moo
1
rodio
The Continental (or International Morse) Code is used for substantially all non-automatic and
of SOUND,
communication. DO NOT memorize from the printed page; code is a language
must not be learned visually; learn by listening as explained in the text.
ceiving tests run for five minutes, and one
minute of errorless transmission or reception
must be accomplished within the five -minute
interval.
If the code test is failed, the applicant must
wait at least one month before he may again
appear for another test. Approximately 30% of
amateur applicants fail to pass the test. It
should be expected that nervousness and excitement will at least to some degree temporarily lower the applicant's code ability. The
best prevention against this is to master the
code at a little greater than the required speed
under ordinary conditions. Then if you slow
down a little due to nervousness during a test
the result will not prove fatal.
There is no shortcut to code pro ficiency. To memorize the alphabet entails but a few evenings of diligent application, but considerable
time is required to build up speed. The exact
time required depends upon the individual's
ability and the regularity of practice.
While the speed of learning will naturally
vary greatly with different individuals, about
70 hours of practice (no practice period to be
over 30 minutes) will usually suffice to bring
a speed of about 13 w.p.m.; 16 w.p.m. requires
about 120 hours; 20 w.p.m., 175 hours.
Memorizing
the Code
Since code reading requires that individual
letters be recognized instantly, any memoriz-
ing scheme which depends upon orderly se-
quence, such as learning all "dab" letters
and all "dit" letters in separate groups, is to
be discouraged. Before beginning with a code
practice set it is necessary to memorize the
whole alphabet perfectly. A good plan is to
study only two or three letters a day and to
drill with those letters until they become part
of your consciousness. Mentally translate each
day's letters into their sound equivalent
wherever they are seen, on signs, in papers,
indoors and outdoors. Tackle two additional
letters in the code chart each day, at the same
time reviewing the characters already learned.
Avoid memorizing by routine. Be able to
sound out any letter immediately without so
much as hesitating to think about the letters
preceding or following the one in question.
Know C, for example, apart from the sequence
ABC. Skip about among all the characters
learned, and before very long sufficient letters
will have been acquired to enable you to spell
out simple words to yourself in "dit dabs."
This is interesting exercise, and for that reason it is good to memorize all the vowels first
and the most common consonants next.
Actual code practice should start only when
the entire alphabet, the numerals, period, corn-
www.americanradiohistory.com
Introduction to Radio
16
THE
RADIO
tion, do it in code. It makes more interesting
practice than confining yourself to random
practice material.
hen two co- learners have memorized the
code and are ready to start sending to each
other for practice, it is a good idea to enlist
the aid of an experienced operator for the first
practice session or two so that they will get
an idea of how properly formed characters
sound.
Figure 2
These code characters are used in languages
other than English. They may occasionally
be encountered so it is well to know them.
ma, and question mark have been memorized
so thoroughly that any one can be sounded
without the slightest hesitation. Do not bother
with other punctuation or miscellaneous signals until later.
-
Each letter and figure must be
memorized by its sound rather
than its appearance. Code is a
system of sound communication, the same as
is the spoken word. The letter A, for example,
is one short and one long sound in combination sounding like dit dab, and it must be remembered as such, and not as "dot dash."
Sound
Not Sight
Practice
Time, patience, and regularity are
required to learn the code properly.
Do not expect to accomplish it within a few
days.
Don't practice too long at one stretch; it
does more harm than good. Thirty minutes at
a time should be the limit.
Lack of regularity in practice is the most
common cause of lack of progress. Irregular
practice is very little better than no practice
at
all. Write down what you have heard; then
forget it; do not look back. If your mind dwells
even for an instant on a signal about which
you have doubt, you will miss the next few
characters while your attention is diverted.
While various automatic code machines,
phonograph records, etc., will give you practice, by far the best practice is to obtain a
study companion who is also interested in
learning the code. When you have both memorized the alphabet you can start sending to
each other. Practice with a key and oscillator
or key and buzzer generally proves superior
to all automatic equipment. Two such sets
operated between two rooms are fine -or between your house and his will be just that
much better. Avoid talking to your partner
while practicing. If you must ask him a ques-
During the first practice period the speed
should be such that substantially solid copy
can be made without strain. Never mind if this
is only two or three words per minute. In the
next period the speed should be increased
slightly to a point where nearly all of the
characters can be caught only through conscious effort. When the student becomes proficient at this new speed, another slight increase may be made, progressing in this manner until a speed of about 16 words per minute
is attained if the object is to pass the amateur
13 -word per minute code test. The margin of
3 w.p.m. is recommended to overcome a possible excitement factor at examination time.
Then when you take the test you don't have to
worry about the "jitters" or an "off day."
Speed should not be increased to a new
level until the student finally makes solid
copy with ease for at least a five -minute
period at the old level. How frequently increases of speed can be made depends upon
individual ability and the amount of practice.
Each increase is apt to prove disconcerting,
but remember "you are never learning when
you are comfortable."
A number of amateurs are sending code
practice on the air on schedule once or twice
each week; excellent practice can be obtained
after you have bought or constructed your re-
ceiver by taking advantage of these sessions.
If you live in a medium -size or large city,
the chances are that there is an amateur radio
club in your vicinity which offers free code
practice lessons periodically.
Skill
listen to someone speaking
you do not consciously think how his
words are spelled. This is also true when you
read. In code you must train your ears to read
code just as your eyes were trained in school
to read printed matter. With enough practice
you acquire skill, and from skill, speed. In
other words, it becomes a habit, something
which can be done without conscious effort.
Conscious effort is fatal to speed; we can't
think rapidly enough; a speed of 25 words a
minute, which is a common one in commercial
operations, means 125 characters per minute
or more than two per second, which leaves
no time for conscious thinking.
When you
www.americanradiohistory.com
Learning the Code
HANDBOOK
Perfect Formation
of Characters
When
transmitting on the
code practice set to your
partner, concentrate on the
quality of your sending, not on your speed.
Your partner will appreciate it and he could
not copy you if you speeded up anyhow.
If you want to get a reputation as having an
excellent "fist" on the air, just remember that
speed alone won't do the trick. Proper execution of your letters and spacing will make
much more of an impression. Fortunately, as
you get so that you can send evenly and accurately, your sending speed will automatically
increase. Remember to try to see how evenly
you can send, and how fast you can receive.
Concentrate on making signals properly with
your key. Perfect formation of characters is
paramount to everything else. Make every signal right no matter if you have to practice it
hundreds or thousands of times. Never allow
yourself to vary the slightest from perfect formation once you have learned it.
If possible, get a good operator to listen to
your sending for a short time, asking him to
criticize even the slightest imperfections.
Timing
It is of the utmost importance to
maintain uniform spacing in characters and combinations of characters. Lack of
uniformity at this point probably causes beginners more trouble than any other single factor. Every dot, every dash, and every space
must be correctly timed. In other words, accurate timing is absolutely essential to intelligibility, and timing of the spaces between
the dots and dashes is just as important as
the lengths of the dots and dashes themselves.
The characters are timed with the dot as a
"yardstick." A standard dash is three times
as long as a dot. The spacing between parts
of the same letter is equal to one dot; the
space between letters is equal to three dots,
and that between words equal to five dots.
The rule for spacing between letters and
words is not strictly observed when sending
slower than about 10 words per minute for the
benefit of someone learning the code and desiring receiving practice. When sending at,
say, 5 w.p.m., the individual letters should be
made the same as if the sending rate were
about 10 w.p.m., except that the spacing between letters and words is greatly exaggerated.
The reason for this is obvious. The letter L,
for instance, will then sound exactly the same
at 10 w.p.m. as at 5 w.p.m., and when the
speed is increased above 5 w.p.m. the student
will not have to become familiar with what
may seem to him like a new sound, although
it is in reality only a faster combination of
dots and dashes. At the greater speed he will
merely have to learn the identification of the
same sound without taking as long to do so.
17
Or-:C,t,.
0oó000boá
taMS
ins
C
B
tmo
tit tt> riti
A
O
Figure
IMP
N
E
3
Diagram illustrating relative lengths of
dashes and spaces referred to the duration
of o dot. A dash is exactly equal in duration
to three dots; spaces between parts of a
letter equal one dot; those between letters,
three dots; space between words, five dots.
Note that a slight increase between two parts
of a letter will make it sound like two
letters.
Be particularly careful of
letters like
B.
Many beginners seem to have a tendency to
leave a longer space after the dash than that
which they place between succeeding dots,
thus making it sound like TS. Similarly, make
sure that you do not leave a longer space after
the first dot in the letter C than you do between other parts of the same letter; otherwise
it will sound like NN.
Once you have memorized the
code thoroughly you should concentrate on increasing your receiving speed. True, if you have to practice
with another newcomer who is learning the
code with you, you will both have to do some
sending. But don't attempt to practice sending
just for the sake of increasing your sending
speed.
When transmitting on the code practice set
to your partner so that he can get receiving
practice, concentrate on the quality of your
sending, not on your speed.
Because it is comparatively easy to learn
to send rapidly, especially when no particular
care is given to the quality of sending, many
operators who have just received their licenses
get on the air and send mediocre or worse code
at 20 w.p.m. when they can barely receive
good code at 13. Most oldtimers remember their
own period of initiation and are only too glad
to be patient and considerate if you tell them
that you are a newcomer. But the surest way
to incur their scorn is to try to impress them
with your "lightning speed," and then to request them to send more slowly when they
come back at you at the same speed.
Stress your copying ability; never stress
your sending ability. It should be obvious that
that if you try to send faster than you can receive, your ear will not recognize any mistakes which your hand may make.
Sending vs.
Receiving
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1
8
I
n
t ro d u c t
i
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t
o
R a d
i
T
o
H E
R A D
I
O
fingers to become tense. Send with a full, free
arm movement. Avoid like the plague any finger motion other than the slight cushioning
effect mentioned above.
Stick to the regular hand key for learning
code. No other key is satisfactory for this purpose. Not until you have thoroughly mastered
both sending and receiving at the maximum
speed in which you are interested should you
tackle any form of automatic or semi -automatic
key such as the Vibroplex ( "bug ") or an electronic key.
Difficulties
Figure 4
PROPER POSITION OF THE FINGERS FOR
OPERATING A TELEGRAPH KEY
The fingers hold the knob and act os a cushion. The hand rests lightly on the key. The
muscles of the forearm provide the power,
the wrist acting as the fulcrum. The power
should not come from the fingers, but rather
from the forearm muscles.
Figure 4 shows the proper position of the hand, fingers and
wrist when manipulating a telegraph or radio
key. The forearm should rest naturally on the
desk. It is preferable that the key be placed
far enough back from the edge of the table
(about 18 inches) that the elbow can rest on
the table. Otherwise, pressure of the table
edge on the arm will tend to hinder the circulation of the blood and weaken the ulnar nerve
at a point where it is close to the surface,
which in turn will tend to increase fatigue
considerably.
The knob of the key is grasped lightly with
the thumb along the edge; the index and third
fingers rest on the top towards the front or far
edge. The hand moves with a free up and down
motion, the wrist acting as a fulcrum. The
power must come entirely from the arm muscles. The third and index fingers will bend
slightly during the sending but not because of
deliberate effort to manipulate the finger muscles. Keep your finger muscles just tight
enough to act as a cushion for the arm motion
and let the slight movement of the fingers take
care of itself. The key's spring is adjusted to
the individual wrist and should be neither too
stiff nor too loose. Use a moderately stiff tension at first and gradually lighten it as you
become more proficient. The separation between the contacts must be the proper amount
for the desired speed, being somewhat under
1/16 inch for slow speeds and slightly closer
together (about 1/32 inch) for faster speeds.
Avoid extremes in either direction.
Do not allow the muscles of arm, wrist, or
Using the Key
Should you experience difficulty
in increasing your code speed
after you have once memorized the characters,
there is no reason to become discouraged. It
is more difficult for some people to learn code
than for others, but there is no justification
for the contention sometimes made that "some
people just can't learn the code." It is not a
matter of intelligence; so don't feel ashamed
if you seem to experience a little more than
the usual difficulty in learning code. Your reaction time may be a little slower or your coordination not so good. If this is the case,
remember you can still learn the code. You
may never learn to send and receive at 40
w.p.m., but you can learn sufficient speed for
all non -commercial purposes and even for most
commercial purposes if you have patience,
and refuse to be discouraged by the fact that
others seem to pick it up more rapidly.
When the sending operator is sending just
a bit too fast for you (the best speed for practice), you will occasionally miss a signal or a
small group of them. When you do, leave a
blank space; do not spend time futilely trying
to recall it; dismiss it, and center attention
on the next letter; otherwise you'll miss more.
Do not ask the sender any questions until the
transmission is finished.
To prevent guessing and get equal practice
on the less common letters, depart occasionally from plain language material and use a jumble of letters in which the usually less commonly used letters predominate.
As mentioned before, many students put a
greater space after the dash in the letter B
than between other parts of the same letter so
it
sounds like TS.
C,
F, Q,
V,
X, Y and Z
often give similar trouble. Make a list of words
or arbitrary combinations in which these letters predominate and practice them, both sending and receiving until they no longer give you
trouble. Stop everything e l s e and stick at
them. So long as they give you trouble you are
not ready for anything else.
Follow the same procedure with letters
which you may tend to confuse such as F and
L, which are often confused by beginners.
www.americanradiohistory.com
HANDBOOK
Figure
Learning the Code
19
5
THE SIMPLEST CODE PRACTICE
SET CONSISTS OF A KEY AND A
INEXPENSIVE 500
OHM POTENTIOMETER
VOLUME CONTROL
BUZZER
is adjusted to give a
steady, high -pitched whine. If desired, the phones may be omitted,
in which case the buzzer should be
mounted firmly on a sounding board.
Crystal, magnetic, or dynamic earphones may be used. Additional
sets of phones should be connected
in parallel, not in series.
The buzzer
=
1.5 TO
S VOLTS
OF BATTERY
1
KEY
Keep at it until you always get them right
without having to stop even an instant to think
about it.
If you do not instantly recognize the sound
of any character, you have not learned it; go
back and practice your alphabet further. You
should never have to omit writing down every
signal you hear except when the transmission
is too fast for you.
Write down what you hear, not what you
think it should be. It is surprising how often
the word which you guess will be wrong.
All good operators copy several words behind, that is,
while one word is being received, they are
writing down or typing, say, the fourth or fifth
previous word. At first this is very difficult,
but after sufficient practice it will be found
actually to be easier than copying close up.
It also results in more accurate copy and enables the receiving operator to capitalize and
Copying Behind
CH-722
COLLECTOR
2= BASE
3= EMITTER
REO
00T
2000 n
PHONES
10K
KEY
0.5 W
Figure
PHONES.
TO 4
PAIR
6
SIMPLE TRANSISTOR CODE
PRACTICE OSCILLATOR
An inexpensive Raytheon CK -722 transistor
requires only a single 11,2 -volt flashlight
battery for power. The inductance of the earphone windings forms part of the oscillatory
circuit. The pitch of the note may be changed
by varying the value of the two capacitors
connected across the earphones.
THESE PARTS REQUIRED
ONLY IF HEADPHONE
OPERATION IS DESIRED
punctuate copy as he goes along. It is not recommended that the beginner attempt to do this
until he can send and receive accurately and
with ease at a speed of at least 12 words a
minute.
It requires a considerable amount of training to dissociate the action of the subconscious mind from the direction of the conscious
mind. It may help some in obtaining this training to write down two columns of short words.
Spell the first word in the first column out loud
while writing down the first word in the second
column. At first this will be a bit awkward,
but you will rapidly gain facility with practice.
Do the same with all the words, and then re-
verse columns.
Next try speaking aloud the words in the one
column while writing those in the other column;
then reverse columns.
After the foregoing can be done easily, try
sending with your key the words in one column while spelling those in the other. It won't
be easy at first, but it is well worth keeping
after if you intend to develop any real code
proficiency. Do not attempt to catch up. There
is a natural tendency to close up the gap, and
you must train yourself to overcome this.
Next have your code companion send you a
word either from a list or from straight text;
do not write it down yet. Now have him send
the next word; after receiving this second
word, write down the first word. After receiving the third word, write the second word; and
so on. Never mind how slowly you must go,
even if it is only two or three words per minute.
Stay behind.
It will probably take quite a number of practice sessions before you can do this with any
facility. After it is relatively easy, then try
staying two words behind; keep this up until
it is easy. Then try three words, four words,
and five words. The more you practice keeping received material in mind, the easier it
will be to stay behind. It will be found easier
at first to copy material with which one is
fairly familiar, then gradually switch to less
familiar material.
www.americanradiohistory.com
20
Introduction
to
R
adio
The two practice sets which
are described in this chapter
are of most value when you
have someone with whom to practice. Automatic code machines are not recommended to anyone who can possibly obtain a companion with
whom to practice, someone who is also interested in learning the code. If you are unable
to enlist a code partner and have to practice
Automatic Code
Machines
by yourself, the best way to g e t receiving
practice is by the use of a tape machine (automatic code sending machine) with several
practice tapes. Or you can use a set of phonograph code practice records. The records are
of use only if you have a phonograph whose
turntable speed is readily adjustable. The tape
machine can be rented by the month for a reasonable fee.
Once you can copy about 10 w.p.m. you can
also get receiving practice by listening to slow
sending stations on your receiver. Many amateur stations send slowly particularly when
working far distant stations. When receiving
conditions are particularly poor many commercial stations also send slowly, sometimes repeating every word. Until you can copy around
10 w.p.m. your receiver isn't much use, and
either another operator or a machine or records
are necessary for getting receiving practice
after you have once memorized the code.
Code Practice
Sets
If you don't feel too foolish
doing it, you can secure a
measure of code practice with
the help of a partner by sending "dit-dah"
messages to each other while riding to work,
eating lunch, etc. It is better, however, to use
a buzzer or code practice oscillator in conjunction with a regular telegraph key.
As a good key may be considered an investment it is wise to make a well -made key your
first purchase. Regardless of what type code
practice set you use, you will need a key, and
later on you will need one to key your trans-
mitter. If you get a good key to begin with,
you won't have to buy another one later.
The key should be rugged and have fairly
heavy contacts. Not only will the key stand
up better, but such a key will contribute to
the "heavy" type of sending so desirable for
radio work. Morse (telegraph) operators use
a "light" style of sending and can send somewhat faster when using this light touch. But,
in radio work static and interference are often
present, and a slightly heavier dot is desirable. If you use a husky key, you will find
yourself automatically sending in this manner.
To generate a tone simulating a code signal
as heard on a receiver, either a mechanical
buzzer or an audio oscillator may be used. Figure 5 shows a simple code-practice set using
a buzzer which may be used directly simply
by mounting the buzzer on a sounding board,
or the buzzer may be used to feed from one to
four pairs of conventional high -impedance
phones.
An example of the audio -oscillator type of
code -practice set is illustrated in figures 6
and 7. An inexpensive Raytheon CK -722 transistor is used in place of the more expensive,
power consuming vacuum tube. A single "pen lite" 1i-volt cell powers the unit. The coils
of the earphones form the inductive portion
of the resonant circuit. 'Phones having an
impedance of 2000 ohms or higher should be
used. Surplus type R -14 earphones also work
well with this circuit.
Figure
7
circuit of Figure 6 is used in this
miniature transistorized code Practice
oscillator. Components are mounted in a
small plastic case. The transistor is
The
attached to a three terminal phenolic
mounting strip. Sub- miniature jacks are
used for the key and phones connections.
A hearing aid earphone may also be used,
as shown. The phone is stored in the
plastic case when not in use.
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CHAPTER TWO
Direct Current Circuits
so different particles, but this further subdivision can be left to quantum mechanics and
atomic physics. As far as the study of electronics is concerned it is only necessary for
the reader to think of the normal atom as being
composed of a nucleus having a net positive
charge that is exactly neutralized by the one
or more orbital electrons surrounding it.
The atoms of different elements differ in
respect to the charge on the positive nucleus
and in the number of electrons revolving
around this charge. They range all the way
from hydrogen, having a net charge of one
on the nucleus and one orbital electron, to
uranium with a net charge of 92 on the nucleus
and 92 orbital electrons. The number of orbital
electrons is called the atomic number of the
element.
All naturally occurring matter (excluding
artifically produced radioactive substances) is
made up of 92 fundamental constituents called
elements. These elements can exist either in
the free state such as iron, oxygen, carbon,
copper, tungsten, and aluminum, or in chemical unions commonly called compounds. The
smallest unit which still retains all the original characteristics of an element is the atom.
Combinations of atoms, or subdivisions of
compounds, result in another fundamental
unit, the molecule. The molecule is the smallest unit of any compound. All reactive elements when in the gaseous state also exist
in the molecular form, made up of two or more
atoms. The nonreactive gaseous elements
helium, neon, argon, krypton, xenon, and
radon are the only gaseous elements that ever
exist in a stable monatomic state at ordinary
temperatures.
From the above it must not be
thought that the electrons revolve in a haphazard manner
around the nucleus. Rather, the electrons in
an element having a large atomic number are
grouped into rings having a definite number of
electrons. The only atoms in which these rings
Action of the
Electrons
The Atom
2-1
An atom is an extremely small unit of
matter there are literally billions of them
making up so small a piece of material as a
speck of dust. To understand the basic theory
of electricity and hence of radio, we must go
further and divide the atom into its main
components, a positively charged nucleus and
a cloud of negatively charged particles that
surround the nucleus. These particles, swirling
around the nucleus in elliptical orbits at an
incredible rate of speed, are called orbital
-
are completely filled are those of the inert
gases mentioned before; all other elements
have one or more uncompleted rings of electrons. If the uncompleted ring is nearly empty,
the element is metallic in character, being
most metallic when there is only one electron
in the outer ring. If the incomplete ring lacks
only one or two electrons, the element is
usually non- metallic. Elements with a ring
about half completed will exhibit both nonmetallic and metallic characteristics; carbon,
silicon, germanium, and arsenic are examples.
Such elements are called semi- conductors.
In metallic elements these outer ring electrons are rather loosely held. Consequently,
electrons.
It is upon the behavior of these electrons
when freed from the atom, that depends the
study of electricity and radio, as well as
allied sciences. Actually it is possible to subdivide the nucleus of the atom into a dozen or
21
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22
Direct Current Circuits
there is a continuous helter -skelter movement
of these electrons and a continual shifting
from one atom to another. The electrons which
move about in a substance are called free
electrons, and it is the ability of these electrons to drift from atom to atom which makes
possible the electric current.
If
the free electrons are numerous and loosely held, the
element is a good conductor.
On the other hand, if there are few free electrons, as is the case when the electrons in an
outer ring are tightly held, the element is a
poor conductor. If there are virtually no free
electrons, the element is a good insulator.
Conductors and
Insulators
2-2
Fundamental Electrical
Units and Relationships
Electromotive Force:
Potential Difference
The free electrons in
a conductor move constantly about and change
their position in a haphazard manner. To
produce a drift of electrons or electric current
along a wire it is necessary that there be a
difference in "pressure" or potential between
the two ends of the wire. This potential difference can be produced by connecting a
source of electrical potential to the ends of
the wire.
As will be explained later, there is an excess of electrons at the negative terminal of
a battery and a deficiency of electrons at the
positive terminal, due to chemical action.
When the battery is connected to the wire, the
deficient atoms at the positive terminal attract
free electrons from the wire in order for the
positive terminal to become neutral. The
attracting of electrons continues through the
wire, and finally the excess electrons at
the negative terminal of the battery are attracted by the positively charged atoms at the
end of the wire. Other sources of electrical
potential (in addition to a battery) are: an
electrical generator (dynamo), a thermocouple,
an electrostatic generator (static machine), a
photoelectric cell, and a crystal or piezoelectric generator.
Thus it is seen that a potential difference
is the result of a difference in the number of
electrons between the two (or more) points in
question. The force or pressure due to a
potential difference is termed the electromotive force, usually abbreviated e. m. f. or
E.M.F. It is expressed in units called volts.
It should be noted that for there to be a
potential difference between two bodies or
points it is not necessary that one have a
positive charge and the other a negative
charge. If two bodies each have a negative
THE
RADIO
charge, but one more negative than the other,
the one with the lesser negative charge will
act as though it were positively charged with
respect to the other body. It is the algebraic
potential difference that determines the force
with which electrons are attracted or repulsed,
the potential of the earth being taken as the
zero reference point.
The flow of electrons along a
conductor due to the application
of an electromotive force constitutes an electric current. This drift is in
addition to the irregular movements of the
electrons. However, it must not be thought
that each free electron travels from one end
of the circuit to the other. On the contrary,
each free electron travels only a short distance
before colliding with an atom; this collision
generally knocking off one or more electrons
from the atom, which in turn move a short
distance and collide with other atoms, knocking off other electrons. Thus, in the general
drift of electrons along a wire carrying an
electric current, each electron travels only a
short distance and the excess of electrons at
one end and the deficiency at the other are
balanced by the source of the e.m.f. When this
source is removed the state of normalcy returns; there is still the rapid interchange of
free electrons between atoms, but there is no
general trend or "net movement" in either
one direction or the other.
The Electric
Current
There are two units of measure ment associated with current,
and they are often confused.
The rate of flou of electricity is stated in
amperes. The unit of quantity is the coulomb.
A coulomb is equal to 6.28 x 10" electrons,
and when this quantity of electrons flows by
a given point in every second, a current of
one ampere is said to be flowing. An ampere
is equal to one coulomb per second; a coulomb
is, conversely, equal to one ampere- second.
Thus we see that coulomb indicates amount,
and ampere indicates rate of flow of electric
current.
Older textbooks speak of current flow as
being from the positive terminal of the e.m.f.
source through the conductor to the negative
terminal. Nevertheless, it has long been an
established fact that the current flow in a
metallic conductor is the electronic flow from
the negative terminal of the source of voltage
through the conductor to the positive terminal.
The only exceptions to the electronic direction
of flow occur in gaseous and electrolytic conductors where the flow of positive ions toward
the cathode or negative electrode constitutes
a positive flow in the opposite direction to the
electronic flow. (An ion is an atom, molecule,
Ampere and
Coulomb
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HANDBOOK
Resistance
or particle which either lacks one or more
electrons, or else has an excess of one or
more electrons.)
In radio work the terms "electron flow" and
"current" are becoming accepted as being
synonymous, but the older terminology is still
accepted in the electrical (industrial) field.
Because of the confusion this sometimes
causes, it is often safer to refer to the direction of electron flow rather than to the direction of the "current." Since electron flow
consists actually of a passage of negative
charges, current flow and algebraic electron
flow do pass in the same direction.
The flow of current in a material
depends upon the ease with
which electrons can be detached from the
atoms of the material and upon its molecular
structure. In other words, the easier it is to
detach electrons from the atoms the more free
electrons there will be to contribute to the
flow of current, and the fewer collisions that
occur between free electrons and atoms the
greater will be the total electron flow.
The opposition to a steady electron flow
is called the resistance of a material, and is
one of its physical properties.
The unit of resistance is the ohm. Every
23
TABLE OF RESISTIVITY
'
Material
Aluminum
Bross
Cadmium
Chromium
Copper
Iron
Silver
Zinc
Nichrome
Const
Manganin
Monet
esist vny in
Ohms per
Temp. Coeff. of
resistance per =C
at 20° C.
Circular
Mil -Foot
0.0049
0.003 to 0.007
0.0038
0.00
0.0039
0.006
0.004
0.0035
0.0002
17
45
46
16
10.4
59
9.8
36
650
0.00001
0.00001
0.0019
295
290
255
FIGURE
1
Resistance
substance has a specific resistance, usually
expressed as ohms per mil -foot, which is determined by the material' s molecular structure
and temperature. A mil -foot is a piece of
material one circular mil in area and one foot
long. Another measure of resistivity frequently
used is expressed in the units microhms per
centimeter cube. The resistance of a uniform
length of a given substance is directly proportional to its length and specific resistance,
and inversely proportional to its cross- sectional area. A wire with a certain resistance for a
given length will have twice as much resistance if the length of the wire is doubled. For
a given length, doubling the cross -sectional
area of the wire will halve the resistance,
while doubling the diameter will reduce the
resistance to one fourth. This is true since
the cross -sectional area of a wire varies as
the square of the diameter. The relationship
between the resistance and the linear dimensions of a conductor may be expressed by the
following equation:
R
R =
r =
l=
A =
Conductors and
resistance in ohms
resistivity in Ohms per mil-foot
length of conductor in feet
cross - sectional area in circular mils
In the
molecular structure of
glass,
porcelain, and mica all electrons are tightly held within their orbits and
there are comparatively few free electrons.
This type of substance will conduct an electric current only with great difficulty and is
known as an insulator. An insulator is said to
have a high electrical resistance.
On the other hand, materials that have a
large number of free electrons are known as
conductors. Alost metals, those elements which
have only one or two electrons in their outer
ring, are good conductors. Silver, copper, and
aluminum, in that order, are the best of the
common metals used as conductors and are
said to have the greatest conductivity, or lowest resistance to the flow of an electric
current.
These units are the volt,
Fundamental
Insulators
many materials such as
the ampere, and the ohm.
They were mentioned in the
preceding paragraphs, but were not completely
defined in terms of fixed, known quantities.
The fundamental unit of current, or rate of
flow of electricity is the ampere. A current of
one ampere will deposit silver from a specified solution of silver nitrate at a rate of
1.118 milligrams per second.
Electrical Units
=-rl
A
Where
The resistance also depends upon temperature, increasing with increases in temperature
for most substances (including most metals),
due to increased electron acceleration and
hence a greater number of impacts between
electrons and atoms. However, in the case of
some substances such as carbon and glass the
temperature coefficient is negative and the
resistance decreases as the temperature increases. This is also true of electrolytes. The
temperature may be raised by the external application of heat, or by the flow of the current
itself. In the latter case, the temperature is
raised by the heat generated when the electrons
and atoms collide.
www.americanradiohistory.com
24
Direct Current Circuits
THE
RADIO
1111111111111111
lu
1
Figure
2
TYPICAL RESISTORS
Shown above are various types of resistors used in electronic circuits. The larger units are
power resistors. On the left is a variable power resistor. Three precision -type resistors ore
shown in the tenter with two small composition resistors beneath them. At the right is o
composition -type potentiometer, used for audio circuitry.
The international standard for the ohm is
the resistance offered by a uniform column of
mercury at 0°C., 14.4521 grams in mass, of
constant cross - sectional area and 106.300
centimeters in length. The expression megohm
(1,000,000 ohms) is also sometimes used
when speaking of very large values of resistance.
A volt is the e.m.f. that will produce a current of one ampere through a resistance of
one ohm. The standard of electromotive force
is the Weston cell which at 20 °C. has a
potential of 1.0183 volts across its terminals.
This cell is used only for reference purposes
in a bridge circuit, since only an infinitesimal
-vw-v-
RESISTANCE
Ri
vor
CONDUCTORS
B2
_-
BATTERY
E
Figure
3
SIMPLE SERIES CIRCUITS
At (A) the battery is in series with a single
resistor. At (B) the battery is in series with
two resistors, the resistors themselves being
in series. The arrows indicate the direction of
electron flow.
amount of current may be drawn from it without disturbing its characteristics.
The relationship between the
electromotive force (voltage),
the flow of current (amperes), and the resistance which impedes the flow of current (ohms),
is very clearly expressed in a simple but
highly valuable law known as Ohm's laun.
This law states that the current in amperes is
equal to the voltage in volts divided by the
resistance in ohms. Expressed as an equation:
Ohm's Law
I
=-RE
If the voltage (E) and resistance (R) are
known, the current (I) can be readily found.
If the voltage and current are known, and the
resistance is unknown, the resistance (R) is
E
equal to
.
When the
voltage is the un-
known quantity, it can be found by multiplying I x R. These three equations are all secured
from the original by simple transposition.
The expressions are here repeated for quick
reference:
E
I
=R
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R=-E
I
E = IR
Resistive Circuits
HANDBOOK
Figure 4
SIMPLE PARALLEL
CIRCUIT
The two resistors RI and R2 are said to be in
parallel since the flow of current is offered
two parallel paths. An electron leaving point
A will pass either through R1 or R2, but not
through both, to reach the positive terminal
of the battery. If a large number of lectrons
are considered, the greater number will pass
through whichever of the two resistors has
the lower resistance.
where I is the current in amperes,
R is the resistance in ohms,
E is the electromotive force in volts.
Application of
All electrical circuits fall in-
Ohm's Law
to one of three classes: series
circuits, parallel circuits, and
series -parallel circuits. A series circuit is
one in which the current flows in a single
continuous path and is of the same value at
every point in the circuit (figure 3). In a parallel circuit there are two or more current
paths between two points in the circuit, as
shown in figure 4. Here the current divides at
A, part going through R, and part through R2i
and combines at B to return to the battery.
Figure 5 shows a series -parallel circuit. There
are two paths between points A and B as in
the parallel circuit, and in addition there are
two resistances in series in each branch of
the parallel combination. Two other examples
of series -parallel arrangements appear in figure 6. The way in which the current splits to
flow through the parallel branches is shown by
the arrows.
In every circuit, each of the parts has some
resistance: the batteries or generator, the connecting conductors, and the apparatus itself.
Thus, if each part has some resistance, no
matter how little, and a current is flowing
through it, there will be a voltage drop across
it. In other words, there will be a potential
difference between the two ends of the circuit
element in question. This drop in voltage is
equal to the product of the current and the
resistance, hence it is called the IR drop.
The source of voltage has an internal resistance, and when connected into a circuit
so that current flows, there will be an IR drop
in the source just as in every other part of the
circuit. Thus, if the terminal voltage of the
source could be measured in a way that would
cause no current to flow, it would be found
to be more than the voltage measured when a
current flows by the amount of the IR drop
25
Figure 5
SERIES-PARALLEL
CIRCUIT
In this type of circuit the resistors are arranged in series groups, and these serlesed
groups ore then placed in parallel.
in the source. The voltage measured with no
current flowing is termed the no load voltage;
that measured with current flowing is the load
voltage. It is apparent that a voltage source
having a low internal resistance is most de-
sirable.
The current flowing in a series
circuit is equal to the voltage
impressed divided by the total
resistance across which the voltage is impressed. Since the same current flows through
every part of the circuit, it is merely necessary to add all the individual resistances to
obtain the total resistance. Expressed as a
formula:
Resistances
in Series
+...
+RN .
Riotai =RI +R2 +R,
if the resistances happened to be
all the same value, the total resistance would
be the resistance of one multiplied by the
number of resistors in the circuit.
Of course,
Consider two resistors, one of
100 ohms and one of 10 ohms,
connected in parallel as in figure 4, with a voltage of 10 volts applied
across each resistor, so the current through
each can be easily calculated.
Resistances
in Parallel
E
I= -R
E = 10
volts
I, =
R = 100 ohms
E = 10
volts
R
ohms
10
Total current
10
= 0.1 ampere
100
-= 1.0 ampere
10
I2
=
10
=
I, +
12
= 1.1 ampere
Until it divides at A, the entire current of
1.1 amperes is flowing through the conductor
from the battery to A, and again from B through
the conductor to the battery. Since this is more
current than flows through the smaller resistor
it is evident that the resistance of the parallel
combination must be less than 10 ohms, the
resistance of the smaller resistor. We can find
this value by applying Ohm's law.
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Direct Current Circuits
26
THE
R=-E
RADIO
A
I
E = 10
I
= 1.1
volts
amperes
10
R
=
1.1
9.09 ohms
The resistance of the parallel combination is
9.09 ohms.
Mathematically, we can derive a simple
formula for finding the effective resistance of
two resistors connected in parallel.
This formula is:
-
R
where
R
R,
R2
R, x R,
+R,
is the unknown resistance,
is the resistance of the first resistor,
is the resistance of the second resistor.
If the effective value required is known,
and it is desired to connect one unknown resistor in parallel with one of known value,
the following transposition of the above formula will simplify the problem of obtaining
the unknown value:
R2
where
R, x R
-
R,
-R
is the effective value required,
R, is the known resistor,
R2 is the value of the unknown resistance necessary to give R when
in parallel with R,.
R
The resultant value of placing a number of
unlike resistors in parallel is equal to the reciprocal of the sum of the reciprocals of the
various resistors. This can be expressed as:
R=
1
- +1
R,
1
R,
Figure 6
OTHER COMMON SERIES -PARALLEL
CIRCUITS
R,
-+...
R,
1
+
-
resistors connected in parallel is always
less than the value of the lowest resistance in
more
the combination. It is well to bear this simple
rule in mind, as it will assist greatly in approximating the value of paralleled resistors.
To find the total resistance of
several resistors connected in
series -parallel, it is usually
easiest to apply either the formula for series
resistors or the parallel resistor formula first,
in order to reduce the original arrangement to
a simpler one. For instance, in figure 5 the
series resistors should be added in each
branch, then there will be but two resistors in
parallel to be calculated. Similarly in figure 7,
although here there will be three parallel resistors after adding the series resistors in
each branch. In figure 6B the paralleled resistors should be reduced to the equivalent
series value, and then the series resistance
values can be added.
Resistances in series -parallel can be solved
by combining the series and parallel formulas
into one similar to the following (refer to
figure 7):
Resistors in
Series Parallel
1
R.
R1
The effective value of placing any number
of unlike resistors in parallel can be determined from the above formula. However, it
is commonly used only when there are three
or more resistors under consideration, since
the simplified formula given before is more
convenient when only two resistors are being
used.
From the above, it also follows that when
two or more resistors of the same value are
placed in parallel, the effective resistance of
the paralleled resistors is equal to the value
of one of the resistors divided by the number
of resistors in parallel.
The effective value of resistance of two or
R,+
+--
1
1
R,
R, + R,
1
Rs
+R6+R,
A
voltage divider is exactly what its name implies: a resistor or a series of resistors connected across a source of voltage from which
various lesser values of voltage may be obtained by connection to various points along
the resistor.
A voltage divider serves a most useful purpose in a radio receiver, transmitter or amplifier, because it offers a simple means of
obtaining plate, screen, and bias voltages of
different values from a common power supply
Voltage Dividers
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HANDBOOK
Voltage
Divider
27
BLEEDER CURRENT
i__
FLOWS BETWEEN
POINTS A AND B
EATERNAL
LOAD
Figure 7
ANOTHER TYPE OF
SERIES -PARALLEL CIRCUIT
Figure 8
SIMPLE VOLTAGE DIVIDER
source. It may also be used to obtain very low
voltages of the order of .01 to .001 volt with
a high degree of accuracy, even though a
means of measuring such voltages is lacking.
The procedure for making these measurements
can best be given in the following example.
Assume that an accurately calibrated voltmeter reading from 0 to 150 volts is available,
and that the source of voltage is exactly 100
volts. This 100 volts is then impressed through
a resistance of exactly 1,000 ohms. It will,
then, be found that the voltage along various
points on the resistor, with respect to the
grounded end, is exactly proportional to the
resistance at that point. From Ohm's law, the
current would be 0.1 ampere; this current remains unchanged since the original value of
resistance (1,000 ohms) and the voltage source
(100 volts) are unchanged. Thus, at a 500 ohm point on the resistor (half its entire resistance), the voltage will likewise be halved
or reduced to 50 volts.
The equation (E = I x R) gives the proof:
E = 500 x 0.1 = 50. At the point of 250 ohms
on the resistor, the voltage will be one -fourth
the total value, or 25 volts (E = 250 x 0.1 = 25).
Continuing with this process, a point can be
found where the resistance measures exactly
1 ohm and where the voltage equals 0.1 volt.
It is, therefore, obvious that if the original
source of voltage and the resistance can be
measured, it is a simple matter to predetermine the voltage at any point along the resistor, provided that the current remains constant,
and provided that no current is taken from the
tap -on point unless this current is taken into
consideration.
Proper design of a voltage
divider for any type of radio
equipment is a relatively
simple matter. The first consideration is the
amount of "bleeder current" to be drawn.
In addition, it is also necessary that the desired voltage and the exact current at each tap
on the voltage divider be known.
Figure 8 illustrates the flow of current in a
simple voltage divider and load circuit. The
light arrows indicate the flow of bleeder current, while the heavy arrows indicate the flow
of the load current. The design of a combined
Voltage Divider
Calculations
CIRCUIT
indicate the manner in which the
current flow divides between the voltage divider
itslf and th externo! load circuit.
The arrows
bleeder resistor and voltage divider, such as
is commonly used in radio equipment, is illustrated in the following example:
A power supply delivers 300 volts and is
conservatively rated to supply all needed current for the receiver and still allow a bleeder
current of 10 milliamperes. The following voltages are wanted: 75 volts at 2 milliamperes
for the detector tube, 100 volts at 5 milliamperes for the screens of the tubes, and
250 volts at 20 milliamperes for the plates of
the tubes. The required voltage drop across R,
is 75 volts, across R, 25 volts, across R, 150
volts, and across R, it is 50 volts. These
values are shown in the diagram of figure 9.
The respective current values are also indicated. Apply Ohm's law:
E
R,
75
=
= 7,500 ohms.
01
R,
R, =
R,
E
25
I
012
E
-_
I
-=
150
2,083 ohms.
8,823 ohms.
.017
50
=-E = .037
= 1,351 ohms.
RTotal = 7,500 + 2,083 + 8,823 +
1,351 = 19,757 ohms.
A 20,000 -ohm resistor with three sliding taps
will be of the approximately correct size, and
would ordinarily be used because of the diffi-
culty in securing four separate resistors of the
exact odd values indicated, and because no
adjustment would be possible to compensate
for any slight error in estimating the probable
currents through the various taps.
When the sliders on the resistor once are
set to the proper point, as in the above ex-
www.americanradiohistory.com
Direct Current Circuits
28
THE
-2
10 + 2 +5
+20
AMPS
1M
RI
MA.
50 VOLTS DROP
0V
0MA
A
-2-AMPS
L1,
R2
10 +2
+5 MA
150 VOLTS DROP
AMPi
-
1
300 VOLTS
+2
-1
(-/
10
RADIO
MA.
25 VOLTS DROP
Figure 10
ILLUSTRATING KIRCHHOFF'S
l
R CURRENT10 .A.J
BLEEDE75
VOLTS D,ROP
POWER SUPPLY
Figure
-
LOA
D
-
-
-
-
-
9
MORE COMPLEX VOLTAGE DIVIDER
The method for computing the values of the
resistors is discussed in the accompanying text.
FIRST LAW
The current flowing toward point "A" is
to the current flowing away from point
qual
"A."
-
sum of all currents flowing toward
and away from the point
taking signs into
account
is equal to zero. Such a sum is
known as an algebraic sum; such that the law
can be stated thus: The algebraic sum of all
tive, the
-
ample, the voltages will remain constant at
the values shown as long as the current remains a constant value.
currents entering and leaving a point is zero.
Figure 10 illustrates this first law. Since the
effective resistance of the network of resistors
is 5 ohms, it can be seen that 4 amperes flow
One of the serious disadvanrages of the voltage divider
becomes evident when the
the current drawn fromone of the taps changes.
It is obvious that the voltage drops are interdependent and, in turn, the individual drops
are in proportion to the current which flows
through the respective sections of the divider
resistor. The only remedy lies in providing a
heavy steady bleeder current in order to make
the individual currents so small a part of the
total current that any change in current will
toward point A, and 2 amperes flow away
through the two 5 -ohm resistors in series. The
remaining 2 amperes flow away through the 10ohm resistor. Thus, there are 4 amperes flowing
to point A and 4 amperes flowing away from
the point. If R is the effective resistance of
the network (5 ohms), R, = 10 ohms, R, = 5
ohms, R, = 5 ohms, and E = 20 volts, we can
set up the following equation:
Disadvantages of
Voltage Dividers
result in only
a
slightchange in voltage. This
can seldom be realized in practice because of
the excessive values of bleeder current which
would be required.
Kirchhoff's Laws
Ohm's law is all that is
necessary to calculate the
values in simple circuits, such as the preceding examples; but in more complex problems, involving several loops or more than
one voltage in the same closed circuit, the
use of Kirchhoff's laws will greatly simplify
the calculations. These laws are merely rules
for applying Ohm's law.
Kirchhoff's first law is concerned with net
current to a point in a circuit and states that:
At any point in a circuit the current
flowing toward the point is equal to
the current flowing away from the
point.
Stated in another way: if currents flowing to
the point are considered positive, and those
flowing from the point are considered nega-
E
E
R
R,
E
20
20
5
10
R2
+R,
=0
20
5
+5
4 -2 -2 =0
Kirchhoff's second law is concerned with
net voltage drop around a closed loop in
a
circuit and states that:
In any closed path or loop in a circuit
the sum of the IR drops must equal
the sum of the applied e.m. f.'s.
The second law also may be conveniently
stated in terms of an algebraic sum as: The
algebraic sum of all voltage drops around a
closed path or loop in a circuit is zero. The
applied e.m.f.'s (voltages) are considered
positive, while IR drops taken in the direction
of current flow (including the internal drop
of the sources of voltage) are considered
negative.
Figure 11 shows an example of the applica-
tion of Kirchhoff's laws to a comparatively
simple circuit consisting of three resistors and
www.americanradiohistory.com
Kirchoff's Laws
HANDBOOK
29
1 volt forces a current of 1 ampere
through a circuit. The power in a resistive
circuit is equal to the product of the voltage applied across, and the current flowing
in, a given circuit. Hence: P (watts) = E
(volts) x I (amperes).
Since it is often convenient to express
power in terms of the resistance of the circuit
and the current flowing through it, a substitution of IR for E (E = IR) in the above formula
gives: P = IR x I or P = 12R. In terms of voltage and resistance, P = E' /R. Here, I = E/R
and when this is substituted for I the original
formula becomes P = E x E /R, or P = E' /R.
To repeat these three expressions:
an e.m.f. of
1.
2.
SET VOLTAGE DROPS AROUND EACH LOOP EQUAL TO ZERO.
1121DHMS)+2(t
-12)+3 =0
-6+2 (12-11)
+312 °0
(FIRST LOOP)
(SECOND LOOP)
SIMPLIFY
211+211-212+3.0
411
+3
-
2
1
21a- 2It+31z -6 =0
512- 211 -6 =0
2
211+6
5
3.
411
+3
2
4
5
-
I
z
P =
EQUATE
2It +6
-
5
SIMPLIFY
and
2011+15= 411 +12
11 ß-t6 AMPERE
I
RE- SUBSTITUTE
Iz-
23
2
EI, P = I2R, and
P = E2 /R,
where P is the power in watts,
E is the electromotive force in volts,
t
2
I
á
AMPERE
Figure 11
ILLUSTRATING KIRCHHOFF'S
SECOND LAW
The voltage drop around any closed loop In a
network Is qual to zero.
two batteries. First assume an arbitrary direction of current flow in each closed loop of the
circuit, drawing an arrow to indicate the assumed direction of current flow. Then equate
the sum of all IR drops plus battery drops
around each loop to zero. You will need one
equation for each unknown to be determined.
Then solve the equations for the unknown currents in the general manner indicated in figure
11. If the answer comes out positive the direction of current flow you originally assumed
was correct. If the answer comes out negative,
the current flow is in the opposite direction to
the arrow which was drawn originally. This is
illustrated in the example of figure 11 where
the direction of flow of I, is opposite to the
direction assumed in the sketch.
In order to cause electrons
Resistive Circuits to flow through a conductor,
constituting a current flow,
it is necessary to apply an electromotive force
(voltage) across the circuit. Less power is
expended in creating a small current flow
through a given resistance than in creating
a large one; so it is necessary to have a unit
of power as a reference.
The unit of electrical power is the watt,
which is the rate of energy consumption when
is the current in amperes.
To apply the above equations to a typical
problem: The voltage drop across a cathode
resistor in a power amplifier stage is 50 volts;
the plate current flowing through the resistor
is 150 milliamperes. The number of watts the
resistor will be required to dissipate is found
from the formula: P = El, or 50 x .150 = 7.5
watts (.150 amperes is equal to 150 milliamperes). From the foregoing it is seen that
a 7.5 -watt resistor will safely carry the required current, yet a 10- or 20 -watt resistor
would ordinarily be used to provide a safety
factor.
In another problem, the conditions being
similar to those above, but with the resistance
(R = 333`/2 ohms), and current being the known
factors, the solution is obtained as follows:
P = I2R = .0225 x 333.33 = 7.5. If only the voltage and resistance are known, P = E2 /R =
2500/333.33 = 7.5 watts. It is seen that all
three equations give the same results; the
selection of the particular equation depends
only upon the known factors.
It
is important to remember
that power (expressed in watts,
horsepower, etc.), represents
the rate of energy consumption or the rate of
doing work. But when we pay our electric bill
Power, Energy
and Work
Power in
Figure 12
MATCHING OF
RESISTANCES
RL
I
To deliver the greatest amount of power to the
load, the load resistance RL should be equal to
the Internal reslstonce of the battery RI.
www.americanradiohistory.com
30
Direct Current Circuits
THE
RADIO
is said to have a certain capacitance. The
energy stored in an electrostatic field is expressed in joules (watt seconds) and is equal
to CE' /2, where C is the capacitance in farads
(a unit of capacitance to be discussed) and E
is the potential in volts. The charge is equal
to CE, the charge being expressed in coulombs.
metallic plates separated from each other by
a thin layer of insulating
material (called a dielectric, in this case),
becomes a capacitor. When a source of d-c
potential is momentarily applied across these
plates, they may be said to become charged.
If the same two plates are then joined together momentarily by means of a switch, the
capacitor will discharge.
When the potential was first applied, electrons immediately flowed from one plate to the
other through the battery or such source of
d -c potential as was applied to the capacitor
plates. However, the circuit from plate to
plate in the capacitor was incomplete (the two
plates being separated by an insulator) and
thus the electron flow ceased, meanwhile establishing a shortage of electrons on one plate
and a surplus of electrons on the other.
Remember that when a deficiency of electrons exists at one end of a conductor, there
is always a tendency for the electrons to move
about in such a manner as to re- establish a
state of balance. In the case of the capacitor
herein discussed, the surplus quantity of electrons on one of the capacitor plates cannot
move to the other plate because the circuit
has been broken; that is, the battery or d -c potential was removed. This leaves the capacitor in a charged condition; the capacitor plate
with the electron deficiency is positively
charged, the other plate being negative.
In this condition, a considerable stress
exists in the insulating material (dielectric)
which separates the two capacitor plates, due
to the mutual attraction of two unlike potentials on the plates. This stress is known as
electrostatic energy, as contrasted with electromagnetic energy in the case of an inductor.
This charge can also be called potential
energy because it is capable of performing
work when the charge is released through an
external circuit. The charge is proportional to
the voltage but the energy is proportional to
the voltage squared, as shown in the following
Capacitance and
Capacitors
sm.
nErg
Figure 13
TYPICAL CAPACITORS
large units ore high value filter capaci-
The two
tors. Shown beneath these ore various types of
by -pass capacitors for r-f and audio application.
to the power company we have purchased a
specific amount of energy or work expressed
in the common units of kilowatt- hours. Thus
rate of energy consumption (watts or kilowatts)
multiplied by time (seconds, minutes or hours)
gives us total energy or work. Other units of
energy are the watt- second, BTU, calorie, erg,
and joule.
Heating Effect
Heat is generated when a
source of voltage causes a
current to flow through a resistor (or, for that
matter, through any conductor). As explained
earlier, this is due to the fact that heat is
given off when free electrons collide with the
atoms of the material. More heat is generated
in high resistance materials than in those of
low resistance, since the free electrons must
strike the atoms harder to knock off other
electrons. As the heating effect is a function
of the current flowing and the resistance of
the circuit, the power expended in heat is
given by the second formula: P = I'R.
2 -3
Electrostatics
-
Capacitors
Electrical energy can be stored in an electrostatic field. A device capable of storing
energy in such a field is called capacitor
(in earlier usage the term condenser was
frequently used but the IRE standards call for
the use of capacitor instead of condenser) and
Two
analogy.
The charge represents a definite amount of
electricity, or a given number of electrons.
The potential energy possessed by these
electrons depends not only upon their number,
but also upon their potential or voltage.
Compare the electrons to water, and two
capacitors to standpipes, a
www.americanradiohistory.com
1
fifd.
capacitor to
Capacitance
HANDBOOK
ROSTATIC
A- EILECT
ELD
-
SHORTAGE
OF ELECTRONS
1
SURPLUS
OF ELECTRONS
1
Figure 14
SIMPLE CAPACITOR
Illustrating the imaginary lines of force repre
Renting the paths along which the repelling force
If the external circuit of
the two capacitor plates is
completed by joining the
terminals together with a piece of wire, the
electrons will rush immediately from one plate
to the other through the external circuit and
establish a state of equilibrium. This latter
phenomenon explains the discharge of a capacitor. The amount of stored energy in a charged
capacitor is dependent upon the charging potential, as well as a factor which takes into
account the size of the plates, dielectric
thickness, nature of the dielectric, and the
number of plates. This factor, which is determined by the foregoing, is called the capacitanceof a capacitor and is expressed in farads.
The farad is such a large unit of capacitance that it is rarely used in radio calculations, and the following more practical units
The Unit of Capacitance: The Farad
have, therefore, been chosen.
farad,
-
CxE'
2 x
1,000,000
This storage of energy in a capacitor is one
of its very important properties, particularly
in those capacitors which are used in power
supply filter circuits.
a standpipe having a cross section of 1 square
inch and a 2 pfd. capacitor to a standpipe having a cross section of 2 square inches. The
charge will represent a given volume of water,
as the "charge" simply indicates a certain
number of electrons. Suppose the water is
equal to 5 gallons.
Now the potential energy, or capacity for
doing work, of the 5 gallons of water will be
twice as great when confined to the 1 sq. in.
standpipe as when confined to the 2 sq. in.
standpipe. Yet the volume of water, or "charge"
is the same in either case.
Likewise a 1 pfd. capacitor charged to 1000
volts possesses twice as much potential
energy as does a 2 pfd. capacitor charged to
500 volts, though the charge (expressed in
coulombs: Q = CE) is the same in either case.
a
micro-microlarad = one - millionth of one millionth of a farad, or 10'E' farads.
Stored energy in joules
of the electrons would act on o free electron
located between the two capacitor plates.
micro farad = 1 /1,000,000 of
.000001 farad, or 10-6 farads.
micro- microfarad = 1 /1,000,000 of a micro farad, or .000001 microfarad, or 10'6 micro farads.
If the capacitance is to be expressed in
microfarads in the equation given for energy
storage, the factor C would then have to be
divided by 1,000,000, thus:
CHARGING CURRENT
1
31
or
Although any substance which has
the characteristics of a good insulator may be used as a dielectric material, commercially manufactured capacitors make use of dielectric materials
which have been selected because their characteristics are particularly suited to the job at
hand. Air is a very good dielectric material,
but an air - spaced capacitor does not have a
high capacitance since the dielectric constant
of air is only slightly greater than one. A
group of other commonly used dielectric mate ials is listed in figure 15.
Certain materials, such as bakelite, lucite,
and other plastics dissipate considerable
energy when used as capacitor dielectrics.
Dielectric
Materials
1
DIELECTRIC
CONSTANT
MATERIAL
1O
ANILINE- FORMALDEHYDE
RESIN
BARIUM TITANATE
MC.
POWER
FACTOR
1O
MC.
3 4
0.004
1200
1.0
.67
CASTOR OIL
3.7
CELLULOSE ACETATE
GLASS.WINDOW
6
GLASS, PYREX
FLUOROTHENE
XEL -F
METHYL - METHACRYLATE
LUCITE
MICA
MYCALEX, MYKROY
PHENOL -FORMALDEHYDE,
LOW-LOSS YELLOW
PHENOL -FORMALDEHYDE
BLACK BAKELITE
PORCELAIN
POLYETHYLENE
POLYSTYRENE
QUARTZ FUSED
RUBBER, HARD-EBONITE
STEATITE
SULFUR
TEFLON
TITANIUM DIOXIDE
TRANSFORMER OIL
UREA -FORMALDEHYDE
VINYL RESINS
WOOD. MAPLE
-
-6
0.04
POOR
SOFTENING
POINT
FAHRENHEIT
260
-
IRO
2000
4.5
0.02
U.S
0.6
-
2.6
5.4
7.0
5.0
0.007
160
5.5
0.03
7.0
25
0.005
0.0003
0.0002
0 0002
0.007
0.003
0.003
0.02
0.0006
0.003
0.05
0.02
2
2.55
4.2
2.6
6.1
3.6
2.1
100 -175
2.2
5.0
4.0
.
FIGURE 15
0.0003
0.002
0.015
POOR
650
270
330
_2600
220
175'
2600
150
2700'
236
-
2700
260
200
Direct Current Circuits
34
1
C
1
1
1
C,
1
1
1
1
C,
C,
C,
C,
capacitor is connected into a direct -current circuit, it will block
the d.c., or stop the flow of current. Beyond
the initial movement of electrons during the
period when the capacitor is being charged,
there will be no flow of current because the
circuit is effectively broken by the dielectric
of the capacitor.
Strictly speaking, a very small current may
actually flow because the dielectric of the
capacitor may not be a perfect insulator. This
minute current flow is the leakage current
previously referred to and is dependent upon
the internal d -c resistance of the capacitor.
This leakage current is usually quite noticeable in most types of electrolytic capacitors.
When an alternating current is applied to
a capacitor, the capacitor will charge and discharge a certain number of times per second
in accordance with the frequency of the alternating voltage. The electron flow in the charge
and discharge of a capacitor when an a-c
potential is applied constitutes an alternating
current, in effect. It is for this reason that a
capacitor will pass an alternating current yet
offer practically infinite opposition to a direct
current. These two properties are repeatedly
in evidence in a radio circuit.
Capacitors in
and
D -C
EQUAL
RESISTANCE
EQUAL
CAPACITANCE
When a
A -C
Circuits
good paper dielectric
filter capacitor has such a
high internal resistance (inin Series
dicating a good dielectric)
that the exact resistance will vary considerably from capacitor to capacitor even though
they are made by the same manufacturer and
are of the same rating. Thus, when 1000 volts
d.c. is connected across two 1-pfd. 500-volt
capacitors in series, the chances are that the
voltage will divide unevenly and one capacitor
will receive more than 500 volts and the other
less than 500 volts.
Voltage Rating
of Capacitors
RADIO
1
- +- - +- - +C,
THE
Any
connecting a half 1 -watt carbon resistor across each capacitor, the voltage will be equalized because the
resistors act as a voltage divider, and the
internal resistances of the capacitors are so
much higher (many megohms) that they have
but little effect in disturbing the voltage divider balance.
Carbon resistors of the inexpensive type
are not particularly accurate (not being designed for precision service); therefore it is
Voltage Equalizing
Resistors
By
megohm
Figure
18
SHOWING THE USE OF VOLTAGE EQUALIZING RESISTORS ACROSS CAPACITORS
CONNECTED IN SERIES
advisable to check several
ohmmeter to find two that
possible in resistance. The
is unimportant, just so it is
two resistors used.
on an accurate
are as close as
exact resistance
the same for the
capacitors are connected in series, alternating
voltage pays no heed to the
relatively high internal resistance of each
capacitor, but divides across the capacitors
in inverse proportion to the capacitance. Because, in addition to the d.c. across a capacitor in a filter or audio amplifier circuit there
is usually an a -c or a -f voltage component, it
is inadvisable to series -connect capacitors
of unequal capacitance even if dividers are
provided to keep the d.c. within the ratings of
the individual capacitors.
For instance, if a 500 -volt 1 -µfd. capacitor
is used in series with a 4-pfd. 500 -volt capacitor across a 250 -volt a -c supply, the 1 -µfd.
capacitor will have 200 volts a.c. across it
and the 4-pfd. capacitor only 50 volts. An
equalizing divider to do any good in this case
would have to be of very low resistance because of the comparatively low impedance of
the capacitors to a.c. Such a divider would
draw excessive current and be impracticable.
The safest rule to follow is to use only
capacitors of the same capacitance and voltage rating and to install matched high resistance proportioning resistors across the various
capacitors to equalize the d-c voltage drop
across each capacitor. This holds regardless
of how many capacitors are series -connected.
Capacitors in
Series on A.C.
When two
Electrolytic capacitors use a very
thin film of oxide as the dielectric, and are polarized; that is,
they have a positive and a negative terminal
which must be properly connected in a circuit;
otherwise, the oxide will break down and the
capacitor will overheat. The unit then will no
longer be of service. When electrolytic capacitors are connected in series, the positive terminal is always connected to the positive lead
of the power supply; the negative terminal of
Electrolytic
Capacitors
HANDBOOK
M
the capacitor connects to the positive terminal
of the next capacitor in the series combination.
The method of connection for electrolytic capacitors in series is shown in figure 18. Electrolytic capacitors have very low cost per
microfarad of capacity, but also have a large
power factor and high leakage; both dependent
upon applied voltage, temperature and the age
of the capacitor. The modern electrolytic capacitor uses a dry paste electrolyte embedded
in a gauze or paper dielectric. Aluminium foil
and the dielectric are wrapped in a circular
bundle and are mounted in a cardboard or metal
box. Etched electrodes may be employed to
increase the effective anode area, and the
total capacity of the unit.
The capacity of an electrolytic capacitor is
affected by the applied voltage, the usage of
the capacitor, and the temperature and humidity
of the environment. The capacity usually drops
with the aging of the unit. The leakage current
and power factor increase with age. At high
frequencies the power factor becomes so poor
that the electrolytic capacitor acts as a series
resistance rather than as a capacity.
Magnetism
2 -4
and Electromagnetism
The common bar or horseshoe magnet is
familiar to most people. The magnetic field
which surrounds it causes the magnet to attract other magnetic materials, such as iron
nails or tacks. Exactly the same kind of magnetic field is set up around any conductor
carrying a current, but the field exists only
while the current is flowing.
Magnetic Fields
Before a potential, or voltage, is applied to a con-
ductor there is no external field, because there
is no general movement of the electrons in
one direction. However, the electrons do progressively move along the conductor when an
e.m.f. is applied, the direction of motion depending upon the polarity of the e.m.f. Since
each electron has an electric field about it, the
flow of electrons causes these fields to build
up into a resultant external field which acts in
a plane at right angles to the direction in
which the current is flowing. This field is
known as the magnetic field.
The magnetic field around a current-carrying
conductor is illustrated in figure 19. The
direction of this magnetic field depends entirely upon the direction of electron drift or
current flow in the conductor. When the flow
is toward the observer, the field about the
conductor is clockwise; when the flow is away
from the observer, the field is counter- clockwise. This is easily remembered if the left
hand is clenched, with the thumb outstretched
agnetism
35
ELECTRON DRIFT
.."-SWITCH
Figure
19
LEFT -HAND RULE
Showing the direction of the magnetic lines of
force produced around a conductor carrying an
electric current.
and pointing in the direction of electron flow.
The fingers then indicate the direction of the
magnetic field around the conductor.
Each electron adds its field to the total external magnetic field, so that the greater the
number of electrons moving along the conductor, the stronger will be the resulting field.
One of the fundamental laws of magnetism
is that like poles repel one another and unlike
poles attract one another. This is true of current- carrying conductors as well as of permanent magnets. Thus, if two conductors are placed
side by side and the current in each is flowing
in the same direction, the magnetic fields will
also be in the same direction and will combine
to form a larger and stronger field. If the current flow in adjacent conductors is in opposite
directions, the magnetic fields oppose each
other and tend to cancel.
The magnetic field around a conductor may
be considerably increased in strength by winding the wire into a coil. The field around each
wire then combines with those of the adjacent
turns to form a total field through the coil
which is concentrated along the axis of the
coil and behaves externally in a way similar
to the field of a bar magnet.
If the left hand is held so that the thumb
is outstretched and parallel to the axis of a
coil, with the fingers curled to indicate the
direction of electron flow around the turns of
the coil, the thumb then points in the direction of the north pole of the magnetic field.
The Magnetic
In the magnetic circuit, the
units which correspond to current, voltage, and resistance
in the electrical circuit are flux, magneto motive force, and reluctance.
Circuit
Flux, Flux
Density
is made up of a drift
of electrons, so is a magnetic
field made up of lines of force, and
the total number of lines of force in a given
magnetic circuit is termed the flux. The flux
depends upon the material, cross section, and
length of the magnetic circuit, and it varies
directly as the current flowing in the circuit.
As a current
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Direct Current Circuits
36
The unit of flux is the maxwell, and the symbol is the Greek letter cp (phi).
Flux density is the number of lines of force
per unit area. It is expressed in gauss if the
unit of area is the square centimeter (1 gauss
= 1 line of force per square centimeter), or
in lines per square inch. The symbol for flux
density is B if it is expressed in gausses, or
B if expressed in lines per square Inch.
The force which produces a
flux in a magnetic circuit
is called magnetomotive force.
It is abbreviated m.m.f. and is designated by
the letter F. The unit of magnetomotive force
is the gilbert, which is equivalent to 1.26 x NI,
where N is the number of turns and I is the
current flowing in the circuit in amperes.
The m.m.f. necessary to produce a given
flux density is stated in gilberts per centimeter (oersteds) (H), or in ampere -turns per
inch (H).
magnetomotive
Force
Magnetic reluctance corresponds
to electrical resistance, and is
the property of a material that opposes the
creation of a magnetic flux in the material.
It is expressed in rels, and the symbol is the
letter R. A material has a reluctance of 1 rel
when an m.m.f. of 1 ampere -turn (NI) generates
a flux of 1 line of force in it. Combinations
of reluctances are treated the same as resistances in finding the total effective reluctance. The specific reluctance of any substance is its reluctance per unit volume.
Except for iron and its alloys, most common
materials have a specific reluctance very
nearly the same as that of a vacuum, which,
for all practical purposes, may be considered
the same as the specific reluctance of air.
Reluctance
Ohm's Law for
The relations between flux,
Magnetic Circuits magnetomotive force, and
reluctance are exactly the
same as the relations between current, voltage, and resistance in the electrical circuit.
These can be stated as follows:
F
F
R
THE
duce in air. It may be expressed by the ratio
B/H or B/H. In other words,
B
ç
= flux, F =
B
or
R
H
H
where p is the premeability, B is the flux
density in gausses, B is the flux density in
lines per square inch, H is the m.m.f. in
gilberts per centimeter (oersteds), and H is
the m.m.f. in ampere -turns per inch. These
relations may also be stated as follows:
B
H=-
or
fi
B
H=-,
f
and B=Hit or
B=
Permeability is similar to electric
conductivity. There is, however,
one important difference: the permeability of
magnetic materials is not independent of the
magnetic current (flux) flowing through it,
although electrical conductivity is substantially independent of the electric current in a
wire. When the flux density of a magnetic
conductor has been increased to the saturation
point, a further increase in the magnetizing
force will not produce a corresponding increase in flux density.
Saturation
magnetic circuit
a magnetization
curve may be drawn for a given unit of material. Such a curve is termed a B -H curve, and
may be determined by experiment. When the
current in an iron core coil is first applied,
the relation between the winding current and
the core flux is shown at A -B in figure 20. If
the current is then reduced to zero, reversed,
brought back again to zero and reversed to the
Calculations
To
simplify
calculations,
F=chR
-
MAGNETIZING FORCE
m.m.f., and
R =
H
reluctance.
Permeability expresses the ease
with which a magnetic field may
be set up in a material as compared with the
effort required in the case of air. Iron, for example, has a permeability of around 2000
times that of air, which means that a given
amount of magnetizing effect produced in an
iron core by a current flowing through a coil
of wire will produce 2000 times the flux density
that the same magnetizing effect would pro-
Hµ
It can be seen from the foregoing that permeability is inversely proportional to the
specific reluctance of a material.
R
where
RADIO
Permeability
Figure
20
TYPICAL HYSTERESIS LOOP
(B -H CURVE = A -B)
Showing relationship between the current in the
winding of on iron core inductor and the core
Inducflux. A direct current flowing through
tance brings the magnetic state of the core to
some point on the hysteresis loop, such as C.
www.americanradiohistory.com
th
Inductance
HANDBOOK
original direction, the flux passes through a
typical hysteresis loop as shown.
The magnetism remaining
in a material after the
magnetizing force is removed is called residual magnetism. Retentivity is the property which causes a magnetic
material to have residual magnetism after
having been magnetized.
Residual Magnetism;
Retentivity
Hysteresis;
Coercive Force
Hysteresis is the character -
istic of a magnetic system
which causes a loss of power
due to the fact that a negative magnetizing
force must be applied to reduce the residual
magnetism to zero. This negative force is
termed coercive /orce. By "negative" magnetizing force is meant one which is of the
opposite polarity with respect to the original
magnetizing force. Hysteresis loss is apparent
in transformers and chokes by the heating of
the core.
the current. Thus, it can be seen that selfinduction tends to prevent any change in the
current in the circuit.
The storage of energy in a magnetic field
is expressed in joules and is equal to (LI3) /2.
(A joule is equal to 1 watt- second. L is defined immediately following.)
Inductance is usually denoted by
the letter L, and is expressed in
henrys. A coil has an inductance
of 1 henry when a voltage of 1
volt is induced by a current change of 1 ampere per second. The henry, while commonly
used in audio frequency circuits, is too large
for reference to inductance coils, such as
those used in radio frequency circuits; millihenry or microhenry is more commonly used,
in the following manner:
The Unit of
Inductance;
The Henry
1
1
If the switch shown in figure 19
is opened and closed, a pulsating
direct current will be produced. When it is
first closed, the current does not instantaneously rise to its maximum value, but builds
up to it. While it is building up, the magnetic
field is expanding around the conductor. Of
course, this happens in a small fraction of a
second. If the switch is then opened. the current stops and the magnetic field contracts
quickly. This expanding and contracting field
will induce a current in any other conductor
that is part of a continuous circuit which it
cuts. Such a field can be obtained in the way
just mentioned by means of a vibrator inter ruptor, or by applying a.c. to the circuit in
place of the battery. Varying the resistance of
the circuit will also produce the same effect.
This inducing of a current in a conductor due
to a varying current in another conductor not
in acutal contact is called electromagnetic induction.
Inductance
Self -inductance
If an alternating current flows
through a coil the varying
magnetic field around each turn cuts itself and
the adjacent turn and induces a voltage in the
coil of opposite polarity to the applied e.m.f.
The amount of induced voltage depends upon
the number of turns in the coil, the current
flowing in the coil, and the number of lines
of force threading the coil. The voltage so
induced is known as a counter-e.m. f. or back e.m.f., and the effect is termed self -induction.
When the applied voltage is building up, the
counter- e.m.f. opposes the rise; when the applied voltage is decreasing, the counter- e.m.f.
is of the same polarity and tends to maintain
37
henry = 1,000
henrys.
1
or
10'
milli -
millihenry = 1 /1,000 of a henry, .001 henry,
or
1
millihenrys,
10'
henry.
microhenry = 1 /1,000,000 of a henry,
.000001 henry, or 10-e henry.
or
microhenry =1/1,000 of a millihenry, .001
or 10-' millibenrys.
1,000 microbenrys =
millihenry.
1
coil is near another, a varying current in
one will produce a varying magnetic field
which cuts the turns of the other coil, inducing
a current in it. This induced current is also
varying, and will therefore induce another current in the first coil. This reaction between
two coupled circuits is called mutual induction,
and can be calculated and expressed in henrys.
The symbol for mutual inductance is M. Two
circuits thus joined are said to be inductively
coupled.
The magnitude of the mutual inductance depends upon the shape and size of the two circuits, their positions and distances apart, and
the premeability of the medium. The extent to
When one
Mutual Inductance
i
I
i.,
u
I
I
2
I
Figure 21
MUTUAL INDUCTANCE
The quantity M represents the mutual inductance
between the two coils L1 and L,.
www.americanradiohistory.com
Direct Current Circuits
38
i--
L
---¡
INDUCTANCE OF
SINGLE- LAYER
SOLENOID COILS
R2 N2
9R +10 L
L
WHERE
R
=
L
=
RADIUS OF COIL
LENGTH OF COIL
N
=
NUMBER OF TURNS
MICRONENRIES
TO CENTER OF
WIRE
Figure 22
FORMULA FOR
CALCULATING INDUCTANCE
Through the usa of the equation and the sketch
shown above
inductance of single -layer
solenoid coils can be calculated with an accuracy of about on. per cent for tho types of
coils normally used in the h -f and v -h -f range.
th
which two inductors are coupled is expressed
by a relation known as coefficient of coupling.
This is the ratio of the mutual inductance actually present to the maximum possible value.
The formula for mutual inductance is L
L, + L, + 2M when the coils are poled so that
their fields add. When they are poled so that
their fields buck, then L = L, + L, - 2M
(figure 21).
Inductors in parallel are corn bined exactly as are resistors in
parallel, provided that they are
far enough apart so that the mutual inductance
is entirely negligible.
Inductors in
Parallel
Inductors in series are additive,
just as are resistors in series,
again provided that no mutual
inductance exists. In this case, the total inductance L is:
Inductors in
Series
L =
L,
etc.
+ L2 +
Where mutual inductance does exist:
L
=L, +L,+
M is the mutual inductance.
This latter expression assumes that the
coils are connected in such a way that all flux
linkages are in the same direction, i.e., additive. If this is not the case and the mutual
linkages subtract from the self -linkages, the
following formula holds:
L
M
=L,
+L,- 2M,
as the frequency is increased. The principal
use for conventional magnetic cores is in the
audio -frequency range below approximately
15,000 cycles, whereas at very low frequencies
(50 to 60 cycles) their use is mandatory if
an appreciable value of inductance is desired.
An air core inductor of only 1 henry inductance would be quite large in size, yet
values as high as 500 henrys are commonly
available in small iron core chokes. The inductance of a coil with a magnetic core will
vary with the amount of current (both a-c and
d-c) which passes through the coil. For this
reason, iron core chokes that are used in power
supplies have a certain inductance rating at a
predetermined value of d-c.
The premeability of air does not change
with flux density; so the inductance of iron
core coils often is made less dependent upon
flux density by making part of the magnetic
path air, instead of utilizing a closed loop of
iron. This incorporation of an air gap is necessary in many applications of iron core coils,
particularly where the coil carries a considerable d -c component. Because the permeability
of air is so much lower than that of iron, the
air gap need comprise only a small fraction of
the magnetic circuit in order to provide a substantial proportion of the total reluctance.
Iron Cored Inductors
at Radio Frequencies
Iron -core inductors may
be used at radio frequencies if the iron is in a
very finely divided form, as in the case of the
powdered iron cores used in some types of r -f
coils and i -f transformers. These cores are
made of extremely small particles of iron. The
particles are treated with an insulating material so that each particle will be insulated from
the others, and the treated powder is molded
with a binder into cores. Eddy current losses
are greatly reduced, with the result that these
special iron cores are entirely practical in circuits which operate up to 100 Mc. in frequency.
2 -5
and R L
R C
voltage divider may be constructed as
figure 23. Kirchhoff's and Ohm's
Laws hold for such a divider. This circuit is
known as an RC circuit.
A
-
Circuits
Ordinary magnetic cores cannot be used for radio frequencies because the eddy current and hysteresis
losses in the core material becomes enormous
Transients
shown in
Time Constant
RC and RL
is the mutual inductance.
Core Material
RADIO
2M,
where
where
THE
When switch S in figure 23 is
placed in position 1, a volt meter across capacitor C will
indicate the manner in which
the capacitor will become charged through the
resistor R from battery B. If relatively large
values are used for R and C, and if a v -t voltmeter which draws negligible current is used
www.americanradiohistory.com
HANDBOOK
Time
Constant
39
to measure the voltage e, the rate of charge of
the capacitor may actually be plotted with the
aid of a stop watch.
It will be found that the voltage e will begin to rise
rapidly from zero the instant the switch is
closed. Then, as the capacitor begins to
charge, the rate of change of voltage across
the capacitor will be found to decrease, the
charging taking place more and more slowly
as the capacitor voltage e approaches the battery voltage E. Actually, it will be found that
in any given interval a constant percentage of
the remaining difference between e and E
will be delivered to the capacitor as an increase in voltage. A voltage which changes in
this manner is said to increase logarithmically,
or is said to follow an exponential curve.
Voltage Gradient
t001
r
:
60
L<
60
W
V40
<
ti
Wo
rt
20
OTIME
44 100
<`
<
FaA
60
óóZ
t. IN TERMS
OF
TIME CONSTANT
PC'
030 60
Hi
Iug
4.9
40.
ózó
0
óZW
<7
á 7ló1
ñ51
20¡C
- --
0
TIME
Time Constant
-
t, IN TERMS OF TIME
Figure
2
A
mathematical
analysis of
the charging of a capacitor in
this manner would show that the relationship
between the battery voltage E and the voltage
across the capacitor e could be expressed in
the following manner:
13
CONSTANT RC
23
TIME CONSTANT OF AN R -C CIRCUIT
Shown at (A) is the circuit upon which is based
the curves of (B) and (C). (8) shows the rate at
which capacitor C will charge from the instant
at which switch S is placed in position 1. (C)
shows the discharge curve of capacitor C from
the instant at which switch S is placed in
position 3.
e = E (1 _ f
-t /Rc)
where e,E,R, and C have the values discussed
above. f = 2.716 (the base of Naperian or
natural logarithms), and t represents the time
which has elapsed since the closing of the
switch. With t expressed in seconds, R and C
Figure
24
TYPICAL INDUCTANCES
The large inductance is a 1000 -watt transmitting coil. To the right and left of this coil are small r -f
chokes. S
I varieties of low power capability coils are shown below, along with various types of r -f
chokes intended for high- frequency operation.
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Direct Current Circuits
40
R
(INCLUDING D.C. RESISTANCE
OF INDUCTOR
L)
i4cc
means that the voltage across the capacitor will have increased to 63.2 per cent of
the battery voltage in an interval equal to the
time constant or RC product of the circuit.
Then, during the next period equal to the time
constant of the RC combination, the voltage
across the capacitor will have risen to 63.2
per cent of the remaining difference in voltage,
or 86.5 per cent of the applied voltage E.
RL Circuit
TIME
t, IN TERMS OF
TIME CONSTANT
}
Figure 25
In the case of
a series combination
of a resistor and an inductor, as
shown in figure 25, the current through the
combination follows a very similar law to that
given above for the voltage appearing across
the capacitor in an RC series circuit. The
equation for the current through the combination is:
TIME CONSTANT OF AN R -L CIRCUIT
Nota that the time constant for the Increase In
current through an R-L. circuit Is identical to
the
rate of Increase in voltage across the
capacitor In on R -C circuit.
may be expressed in farads and ohms, or R
and C may be expressed in microfarads and
megohms. The product RC is called the time
constant of the circuit, and is expressed in
seconds. As an example, if R is one megohm
and C is one microfarad, the time constant
RC will be equal to the product of the two,
or one second.
When the elapsed time t is equal to the
time constant of the RC network under consideration, the exponent of E becomes -1.
Now
is equal to 1 /e, or 1/2.716, which
is 0.368. The quantity (1 - 0.368) then is equal
to 0.632. Expressed as percentage, the above
e'
i=-E
(1-E-tR/L)
where i represents the current at any instant
through the series circuit, E represents the
applied voltage, and R represents the total
resistance of the resistor and the d-c resistance of the inductor in series. Thus the time
constant of the RL circuit is L /R, with R expressed in ohms and L expressed in henrys.
Voltage Decoy
When the
switch in figure
23
is
moved to position 3 after the
capacitor has been charged, the capacitor voltage will drop in the manner shown in figure
23 -C. In this case the voltage across the capacitor will decrease to 36.8 per cent of the
initial voltage (will make 63.2 per cent of the
total drop) in a period of time equal to the
time constant of the RC circuit.
TYPICAL IRON -CORE INDUCTANCES
At the right is an upright mounting filter choke intended for use in low powered transmitters and audio equipment. At the center is o hermetically sealed inductance for use
under poor environmental conditions. To the left is an inexpensive receiving -type choke,
with a small iron -core r -f choke directly in front of it.
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CHAPTER THREE
Alternating Current Circuits
present the usable frequency range for alternating electrical currents extends over the enormous frequency range from about 15 cycles per
second to perhaps 30,000,000,000 cycles per
second. It is obviously cumbersome to use a
frequency designation in c.p.s. for enormously
high frequencies, so three common units which
are multiples of one cycle per second have
been established.
The previous chapter has been devoted to
discussion of circuits and circuit elements
upon which is impressed a current consisting
of a flow of electrons in one direction. This
type of unidirectional current flow is called
direct current, abbreviated d. c. Equally as important in radio and communications work,
and power practice, is a type of current flow
whose direction of electron flow reverses
periodically. The reversal of flow may take
place at a low rate, in the case of power systems, or it may take place millions of times
per second in the case of communications
frequencies. This type of current flow is
called alternating current, abbreviated a. c.
At
Frequency Spectrum
a
z
4-
Y.1
¢
K
U
3 -1
TIME-41.
a
DIRECT CURRENT
Alternating Current
t CYCLE
Frequency of on
Alternating Current
An
-i
alternating current is
one whose amplitude of
current flow periodically
rises from zero to a maximum in one direction,
decreases to zero, changes its direction,
rises to maximum in the opposite direction,
and decreases to zero again. This complete
process, starting from zero, passing through
two maximums in opposite directions, and returning to zero again, is called a cycle. The
number of times per second that a current
passes through the complete cycle is called
the frequency of the current. One and one
quarter cycles of an alternating current wave
are illustrated diagrammatically in figure 1.
Iz
w
CYCLE
-01
TIME
a
CC
J
U
ALTERNATING CURRENT
Figure
1
ALTERNATING CURRENT
AND DIRECT CURRENT
Graphical comparison between unidrectionai
(direct) current and alternating current as plotted
against time.
41
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-
42
Alternating Current Circuits
THE
RADIO
These units are:
(1) the kilocycle (abbr., kc.), 1000 c.p.s.
(2) the Megacycle (abbr., Mc.), 1,000,000
c.p.s. or 1000 kc.
(3) the kilo -Megacycle (abbr., kN1c.),
1,000,000,000 c.p.s. or 1000 Mc.
easily handled units such as these we
can classify the entire usable frequency range
into frequency bands.
The frequencies falling between about 15
and 20,000 c.p.s. are called audio frequencies,
abbreviated a.f., since these frequencies are
audible to the human ear when converted from
electrical to acoustical signals by a loudspeaker or headphone. Frequencies in the
vicinity of 60 c.p.s. also are called power frequencies, since they are commonly used to
distribute electrical power to the consumer.
The frequencies falling between 10,000
c.p.s. (10 kc.) and 30,000,000.000 c.p.s. (30
kMc.) are commonly called radio frequencies,
abbreviated r. J., since they are commonly used
in radio communication and allied arts. The
radio- frequency spectrum is often arbitrarily
classified into seven frequency bands, each
one of which is ten times as high in frequency
as the one just below it in the spectrum (except for the v -1 -f band at the bottom end of
the spectrum). The present spectrum, with
classifications, is given below.
With
Frequency
kc.
30 to 300 kc.
300 to 3000 kc.
3 to 30 Mc.
30 to 300 Mc.
300 to 3000 Mc.
3 to 30 kMc.
30 to 300 kMc.
10 to 30
Generation of
Alternating Current
Classification
Very -low frequencies
Low frequencies
Medium frequencies
High frequencies
Very -high frequencies
Ultra -high frequencies
Abbrev.
v.l.f
l.f.
m.f.
h. f.
v.h.f.
u.h.f.
Super-high frequencies s.h.f.
Extremely -high
frequencies
e.h.f.
Faraday discovered that if
a conductor which forms
part of a closed circuit is
moved through a magnetic field so as to cut
across the lines of force, a current will flow
in the conductor. He also discovered that, if a
conductor in a second closed circuit is brought
near the first conductor and the current in the
first one is varied, a current will flow in the
second conductor. This effect is known as
induction, and the currents so generated are
induced currents. In the latter case it is the
lines of force which are moving and cutting
the second conductor, due to the varying current strength in the first conductor.
A current is induced in a conductor if there
is a relative motion between the conductor
and a magnetic field, its direction of flow depending upon the direction of the relative
Figure 2
THE ALTERNATOR
Semi -schematic representation of the simplest
form of the alternator.
motion between the conductor and the field,
and its strength depends upon the intensity of
the field, the rate of cutting lines of force, and
the number of turns in the conductor.
machine that generates an alternating current is called an alternator or a -c generator. Such a machine in its
basic form is shown in figure 2. It consists of
two permanent magnets, M. the opposite poles
of which face each other and are machined so
that they have a common radius. Between
these two poles, north (N) and south (S),
a substantially constant magnetic field exists.
If a conductor in the form of C is suspended
so that it can be freely rotated between the
two poles, and if the opposite ends of conductor C are brought to collector rings, there
will be a flow of alternating current when conductor C is rotated. This current will flow out
through the collector rings R and brushes B
to the external circuit, X -Y.
The field intensity between the two pole
pieces is substantially constant over the entire
area of the pole face However, when the
conductor is moving parallel to the lines of
force at the top or bottom of the pole faces,
no lines are being cut. As the conductor moves
on across the pole face it cuts more and more
lines of force for each unit distance of travel,
until it is cutting the maximum number of
lines when opposite the center of the pole.
Therefore, zero current is induced in the conductor at the instant it is midway between
the two poles, and maximum current is induced when it is opposite the center of the
pole face. After the conductor has rotated
through 180° it can be seen that its position
with respect to the pole pieces will be exactly
opposite to that when it started. Hence, the
second 180° of rotation will produce an alternation of current in the opposite direction to
that of the first alternation.
The current does not increase directly as
the angle of rotation, but rather as the sine
of the angle; hence, such a current has the
mathematical form of a sine wave. Although
Alternators
A
www.americanradiohistory.com
HANDBOOK
LINES
Sine /lave
The
43
OF FORCE
t CYCLE
E
le2
90
60
A B C O E
CYCLE
30
HM-
--+
CYCLE- w
t20
ISO
te0
3
a
2
LINES OF FORCE
(UNIFORM DENSITY
240
2t0
3 CYCLE'
Graph showing sine -wave output current of the
alternator of figure
CYCLE' +-
WHERE
F =
TIMES
330
2143
t
Figure 3
OUTPUT OF THE ALTERNATOR
--
300
FREQUENCY IN CYCLES
2.
Figure
most electrical machinery does not produce a
strictly pure sine curve, the departures are
usually so slight that the assumption can be
regarded as fact for most practical purposes.
All that has been said in the foregoing paragraphs concerning alternating current also is
applicable to alternating voltage.
The rotating arrow to the left in figure 3
represents a conductor rotating in a constant
magnetic field of uniform density. The arrow
also can be taken as a vector representing the
strength of the magnetic field. This means that
the length of the arrow is determined by the
strength of the field (number of lines of force),
which is constant. Now if the arrow is rotating
at a constant rate (that is, with constant
angular velocity), then the voltage developed
across the conductor will be proportional to
the rate at which it is cutting lines of force,
which rate is proportional to the vertical
distance between the tip of the arrow and the
horizontal base line.
If EO is taken as unity or a voltage of 1,
then the voltage (vertical distance from tip of
arrow to the horizontal base line) at point C
for instance may be determined simply by
referring to a table of sines and looking up the
sine of the angle which the arrow makes with
the horizontal.
When the arrow has traveled from A to point
E, it has traveled 90 degrees or one quarter
cycle. The other three quadrants are not shown
because their complementary or mirror relationship to the first quadrant is obvious.
It is important to note that time units are
represented by degrees or quadrants. The fact
that AB, BC, CD, and DE are equal chords
(forming equal quadrants) simply means that
the arrow (conductor or vector) is traveling
at a constant speed, because these points on
the radius represent the passage of equal
units of time.
The whole picture can be represented in
another way, and its derivation from the foregoing is shown in figure 3. The time base is
represented by a straight line rather than by
4
THE SINE WAVE
illustrating
one cycle of o sine wave. One
complete cycle of alternation is broken up
into 360 degrees. Then one -half cycle is 180
degrees, one -quarter cycle is 90 degrees, and
so on down to the smallest division of the
wave. A cosine wave has a shape identical to
a sine wave but is shifted 90 degrees in phase
In other words the wove begfna at full am
plilude, the 90- degree point comes at zero amplitude, the 180 -degree point comes at full
amplitude in the opposite direction of current
How, etc.
-
angular rotation. Points A, B, C, etc., represent the same units of time as before. When
the voltage corresponding to each point is
projected to the corresponding time unit, the
familiar sine curve is the result.
The frequency of the generated voltage is
proportional to the speed of rotation of the
alternator, and to the number of magnetic poles
in the field. Alternators may be built to produce
radio frequencies up to 30 kilocycles, and
some such machines are still used for low
frequency communication purposes. By means
of multiple windings, three -phase output may
be obtained from large industrial alternators.
From figure 1 we see that the
value of an a -c wave varies
continuously. It is often of importance to know
the amplitude of the wave in terms of the
total amplitude at any instant or at any time
within the cycle. To be able to establish the
instant in question we must be able to divide
Radian Notation
the cycle into parts. We could divide the cycle
into eighths, hundredths, or any other ratio that
suited our fancy. However, it is much more
convenient mathematically to divide the cycle
either into electrical degrees (360° represent
one cycle) or into radians. A radian is an arc
of a circle equal to the radius of the circle;
hence there are 2n radians per cycle-or per
circle (since there are n diameters per circumference, there are 2rr radii).
Both radian notation and electrical degree
www.americanradiohistory.com
notation are used in discussions of alternating
current circuits. However, trigonometric tables
are much more readily available in terms of
degrees than radians, so the following simple
conversions are useful.
2n radians = 1 cycle = 3600
n radians ='/2 cycle = 180°
- radians ='4 cycle
RADIO
THE
Alternating Current Circuits
44
WHERE
o
e (THETA). PHASE
B
RADIANS
=
A
A
B'/r
D.
t
ANGLE
OR
RADIANS OR
T
.277F T
90'
IRO
RADIANS OR
2 A RADIANS OR
270
350
RADIAN a 57.324 DEGREES
n
=
2
- radians ='4 cycle
90°
Figure
n
=
60°
=
45°
3
-4 radians ='
n
/R
1
radian
cycle
Current
where e
The instantaneous volt age or current is proportional to the sine of the
angle through which the
rotating vector has travelled since reference
time t = 0. Hence, when the peak value of the
a -c wave amplitude (either voltage or current amplitude) is known, and the angle through
which the rotating vector has travelled is
established, the amplitude of the wave at
this instant can be determined through use
of the following expression:
e = Erna: sin
matical relationships involving phase angles
since such relationships are simplified when
radian notation is used
-cycle = 57.3°
When the conductor in the simple alternator of figure 2 has made one complete revolution it has generated one cycle and has rotated through 2n radians. The expression 2nf
then represents the number of radians in one
cycle multiplied by the number of cycles per
second (the frequency) of the alternating
voltage or current. The expression then represents the number of radians per second through
which the conductor has rotated. Hence 27rf
represents the angular velocity of the rotating
conductor, or of the rotating vector which
represents any alternating current or voltage,
expressed in radians per second.
In technical literature the expression 2nf
is often replaced by al, the lower -case Greek
letter omega. Velocity multiplied by time
gives the distance travelled. so 2nft (or an)
represents the angular distance through which
the rotating conductor or the rotating vector
has travelled since the reference time t = 0.
In the case of a sine wave the reference time
t = 0 represents that instant when the voltage
or the current, whichever is under discussion,
also is equal to zero.
Instantaneous Value
The radian is a unit of phase angle, equal to
57.324 degrees. It is commonly used in mathe-
1
=
2n
of Voltage or
5
ILLUSTRATING RADIAN NOTATION
left,
=
the instantaneous voltage
crest value of voltage,
E = maximum
f =
frequency in cycles per second, and
has elapsed
expressed as a fraction
of one second.
t = period of time which
since
t = 0
The instantaneous current can be found from
the same expression by substituting i for e
and Imax for Emaz.
It is often easier to visualize the process of
determining the instantaneous amplitude by
ignoring the frequency and considering only
one cycle of the a -c wave. In this case, for a
sine wave, the expression becomes:
e=Ema: sin 9
where O represents the angle through which
the vector has rotated since time (and amplitude) were zero. As examples:
when 0 = 30°
sin O = 0.5
so e = 0.5 Emu
when
O
=
sin O
so e
=
=
60°
0.866
0.866
Erna:
when
O = 90°
sin 0 = 1.0
so e = Emax
when
sin
so
www.americanradiohistory.com
O
=
=
e =
O
1 radian
0.8415
0.8415 Etna:
H
A-C
A N D B O O K
Effective Value
The instantaneous
of an
of an alternating current
or voltage varies continuously throughout the cycle.
Alternating Current
Relationships
45
value
value of an a -c wave must be chosen
to establish a relationship between the effectiveness of an a -c and a d -c voltage or cur rent/ The heating value of an alternating
current has been chosen to establish the reference between the effective values of a.c. and
d.c. Thus an alternating current will have an
effective value of 1 ampere when it produces
the same heat in a resistor as does 1 ampere
of direct current.
The effective value is derived by taking the
instantaneous values of current over a cycle of
alternating current, squaring these values.
taking an average of the squares, and then
taking the square root of the average. By this
procedure, the effective value becomes known
as the root mean square or r.m.s. value. This
is the value that is read on a -c voltmeters and
a -c ammeters. The r.m.s. value is 70.7 (for
sine waves only) per cent of the peak or maximum instantaneous value and is expressed as
So some
follows:
Eetf. or Er.m.s.
=
0.707 x
left. or Ir.m.s.
=
0.707 x !ma:.
Erna:
or
The following relations are extremely useful
in radio and power work:
Er
m. s. =
Ems
0.707 x
= 1.414 x
&max,
and
Er.m.s.
If
an alternating current
is passed through a rectifier, it emerges in the
form of a current of
varying amplitude which flows in one direction only. Such a current is known as rectified
a. c. or pulsating d. c. A typical wave form of a
pulsating direct current as would be obtained
from the output of a full -wave rectifier is
shown in figure 6.
Measuring instruments designed for d -c
operation will not read the peak for instantaneous maximum value of the pulsating d -c output from the rectifier; they will read only the
average value. This can be explained by assuming that it could be possible to cut off
some of the peaks of the waves, using the cutoff portions to fill in the spaces that are open,
thereby obtaining an average d -c value. A
milliammeter and voltmeter connected to the
adjoining circuit, or across the output of the
rectifier, will read this average value. It is related to peak value by the following expres-
Rectified Alternating
Current or Pulsating Direct Current
sion:
Eavg =
0.636 x Fina:
Figure 6
FULL -WAVE RECTIFIED
SINE WAVE
Waveform obtained at the output of a fullwave
rectifier being fed with a sine wave and having
100 per
cent rectification efficiency. Each
pulse has the same shape os one -half cycle of
a
sine wave. This type of current is known as
pulsating direct current.
It is thus seen that the average value is 63.6
per cent of the peak value.
To summarize the three
most significant values
of an a-c sine wave: the
Effective, and
Average Values
peak value is equal to
1.41 times the r.m.s. or
effective, and the r.m.s. value is equal to
0.707 times the peak value; the average value
of a full -wave rectified a-c wave is 0.636
times the peak value, and the average value
of a rectified wave is equal to 0.9 times the
r.m.s. value.
Relationship Between
Peak, R.M.S. or
= 0.707 x Peak
Average = 0.636 x Peak
R.M.S.
x R.M.S.
Average = 0.9
= 1.11 x Average
R.M.S.
Peak
Peak
= 1.414 x R.M.S.
= 1.57 x Average
law
applies
equally to direct or alternating current, provided the circuits under consideration are
purely resistive, that is, circuits which have
neither inductance (coils) nor capacitance
(capacitors). Problems which involve tube
filaments, drop resistors, electric lamps,
heaters or similar resistive devices can be
solved from Ohm's law, regardless of whether
the current is direct or alternating. When a
capacitor or coil is made a part of the circuit,
a property common to either, called reactance,
must be taken into consideration. Ohm's law
still applies to a -c circuits containing reactance, but additional considerations are involved; these will be discussed in a later
Applying Ohm's Law
to Alternating Current
paragraph.
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Ohm's
Alternating Current Circuits
46
THE
RADIO
E
TIME
TIME
CURRENT LAGGING VOLTAGE BY 90°
CURRENT LEADING VOLTAGE BY 90°
(CIRCUIT CONTAINING PURE INDUCTANCE ONLY)
(CIRCUIT CONTAINING PURE CAPACITANCE ONLY)
Figure 7
LAGGING PHASE ANGLE
Figure 8
LEADING PHASE ANGLE
Showing the manner in which the current lags
the voltage in an a-c circuit containing pure
inductance only. The lag is equal to one -quarter
cycle or 90 degrees.
Inductive
As
Reactance
when
was stated in
Chapter Two,
changing current flows
through an inductor a back- or
counter -electromotive force is developed,
opposing any change in the initial current.
This property of an inductor causes it to offer
opposition or impedance to a change in current. The measure of impedance offered by an
inductor to an alternating current of a given
frequency is known as its inductive reactance.
This is expressed as XL.
a
XL
= 2rrf
L,
n= 3.1416 (2n= 6.283),
f = frequency in cycles,
L = inductance in henrys.
It is very often neces-
cary to compute inductive reactance at radio
frequencies. The same formula may be used,
but to make it less cumbersome the inductance
is expressed in millihenrys and the frequency
in kilocycles. For higher frequencies and
smaller values of inductance, frequency is
expressed in megacycles and inductance in
microhenrys. The basic equation need not be
changed, since the multiplying factors for
inductance and frequency appear in numerator
and denominator, and hence are cancelled out.
However, it is not possible in the same equation to express L in millihenrys and f in cycles
without conversion factors.
Capacitive
Reactance
Capacitors have a similar property although
in this case the opposition is to any change in
the voltage across the capacitor. This property
is called capacitive reactance and is expressed as follows:
Xc
It has been explained that inductive reactance is the measure of
the ability of an inductor to offer
impedance to the flow of an alternating current.
1
2nfC
where Xc = capacitive reactance in ohms,
n = 3.1416
f = frequency in cycles,
C =
where XL = inductive reactance expressed in
ohms.
Inductive Reactance
at Rodio Frequencies
Showing the manner in which the current leads
the voltage in on o -c circuit containing pure
capacitance only. The lead is equal to one quarter cycle or 90 degrees.
capacitance in farads.
Capacitive Reactance at
Radio Frequencies
Here again, as in the case
of inductive reactance, the
units of capacitance and
frequency can be converted
into smaller units for practical problems
en-
countered in radio work. The equation may
be
written:
1,000,000
Xc
2nfC
where f = frequency in megacycles,
C w capacitance in micro- microfarads.
In the audio range it is often convenient to
express frequency (f) in cycles and capacitance (C) in micro /arads, in which event the
same formula applies.
Phase
When
an
alternating current flows
through a purely resistive circuit, it
will be found that the current will go through
maximum and minimum in perfect step with
the voltage. In this case the current is said to
be in step or in phase with the voltage. For
this reason, Ohm's law will apply equally well
for a. c. or d. c. where pure resistances are concerned, provided that the same values of the
www.americanradiohistory.com
Reactance
HANDBOOK
Y-AniS
wave (either peak or r.m.s.) for both voltage
and current are used in the calculations.
However, in calculations involving alternating currents the voltage and current are not
necessarily in phase. The current through the
circuit may lag behind the voltage, in which
case the current is said to have lagging phase.
Lagging phase is caused by inductive reactance. If the current reaches its maximum value
ahead of the voltage (figure 8) the current is
said to have a leading phase. A leading phase
angle is caused by capacitive reactance.
In an electrical circuit containing reactance
only, the current will either lead or lag the
voltage by 90 °. If the circuit contains inductive reactance only, the current will lag the
voltage by 90 °. If only capacitive reactance is
in the circuit, the current will lead the voltage
by 90 °.
Inductive and capacitive reactance have exactly opposite
effects on the phase relation
between current and voltage in a circuit.
Hence when they are used in combination
their effects tend to neutralize. The combined
effect of a capacitive and an inductive reactance is often called the net reactance of a
circuit. The net reactance (X) is found by subtracting the capacitive reactance from the inductive reactance, X = XL Xc.
The result of such a combination of pure
reactances may be either positive, in which
case the positive reactance is greater so that
the net reactance is inductive, or it may be
negative in which case the capacitive reactance is greater so that the net reactance is
capacitive. The net reactance may also be
zero in which case the circuit is said to be
resonant. The condition of resonance will be
discussed in a later section. Note that inductive reactance is always taken as being positive while capacitive reactance is always
taken as being negative.
Reactances
in Combination
Pure reactances introduce a phase angle of
90° between voltage and
and Resistance
current; pure resistance
introduces no phase shift between voltage and
current. Hence we cannot add a reactance and
a resistance directly. When a reactance and a
resistance are used in combination the resulting phase angle of current flow with respect to the impressed voltage lies somewhere
between plus or minus 90° and 0° depending
upon the relative magnitudes of the reactance
and the resistance.
The term impedance is a general term which
can be applied to any electrical entity which
impedes the flow of current. Hence the term
may be used to designate a resistance, a pure
Impedance; Circuits
Containing Reactance
47
Figure
9
Operation on the vector (+A) by the quantity ( -1)
vector to rotate through 180 degrees.
reactance, or a complex combination of both
reactance and resistance. The designation for
impedance is Z. An impedance must be defined in such a manner that both its magnitude
and its phase angle are established. The
designation may be accomplished in either of
two ways-one of which is convertible into
the other by simple mathematical operations.
"J"
The first method of designating an impedance is
actually to specify both the resistive and the
reactive component in the form R + jX. In this
form R represents the resistive component in
ohms and X represents the reactive component.
The "j" merely means that the X component
is reactive and thus cannot be added directly
to the R component. Plus jX means that the
reactance is positive or inductive, while if
minus jX were given it would mean that the
reactive component was negative or capacitive.
In figure 9 we have a vector ( +A) lying along
the positive X-axis of the usual X -Y coordinate system. If this vector is multiplied by the
quantity ( -1), it becomes ( A) and its position
now lies along the X -axis in the negative
direction. The operator ( -1) has caused the
vector to rotate through an angle of 180 deThe
Operator
grees. Since ( -1) is equal to (V-1 x V77-1), the
same result may be obtained by operating on
the vector with the operator (VIT x V-1).
However if the vector is operated on but once
by the operator (V-1), it is caused to rotate
only 90 degrees (figure 10). Thus the operator
( V- 11)rotates a vector by 90 degrees. For convenience, this operator is called the j operator.
rotates the
In like fashion, the operator (
vector of figure 9 through an angle of 270
degrees, so that the resulting vector ( jA)
falls on the ( Y) axis of the coordinate system.
www.americanradiohistory.com
j)
48
Alternating Current Circuits
RADIO
THE
Y-AXIS
(AI
( +A) X
) ROTATES
VECTOR THROUGH 90
+jA
k
4
b
A
X
Z. 4+J3
AXIS
X
i
Figure
(j)
Polar Notation
The second method of representing an impedance is to
specify its absolute magnitude and the phase
angle of current with respect to voltage, in
the form Z L O. Figure 11 shows graphically
the relationship between the two common ways
of representing an impedance.
The construction of figure 11 is called an
impedance diagram. Through the use of such
a diagram we can add graphically a resistance
and a reactance to obtain a value for the resulting impedance in the scalar form. With
zero at the origin, resistances are plotted to
the right, positive values of reactance (inductive) in the upward direction, and negative
values of reactance (capacitive) in the downward direction.
Note that the resistance and reactance are
drawn as the two sides of a right triangle,
with the hypotenuse representing the resulting
impedance. Hence it is possible to determine mathematically the value of a resultant
impedance through the familiar right -triangle
relationship-the square of the hypotenuse is
equal to the sum of the squares of the other
two sides:
=R' +X2
or IZI = ß/R2 + X
Note also that the angle O included between
R and Z can be determined from any of the
following trigonometric relationships:
X
sin o
cos
tan
Z
R
O =
Z
X
O
=
-
IZI'
-R
OHMS
RESISTANCE
R.
5
IZI= 5
t0/-' 0.73
ae.e5
10
Operation on the vector ( +A) by the quantity
causes vector to rotate through 90 degrees.
Z2
o
L 3e.e5
R
One common problem is that of determining
the scalar magnitude of the impedance, IZI,
Figure 11
THE IMPEDANCE TRIANGLE
Showing the graphical construction of a triangle
for obtaining the net (scalar) impedance resulting from the connection of o resistance and
a reactance in series. Shown also alongside is
the
alternative mathematical procedure for
obtaining the values associated with the triangle.
and the phase angle 0, when resistance and
reactance are known; hence, of converting
from the Z = R + jX to the IZI LO form. In this
case we use two of the expressions just given:
IZI = V/R2+X2
tan
-, (or
X
O
=
O =
R
The
tan'
X
)
R
inverse problem,
that of converting
jX form is done
with the following relationships, both of which
are obtainable by simple division from the
trigonometric expressions just given for determining the angle 0:
from the IZI LO to the R +
R
jX
=IZI
cos0
=IZIj
sin
O
By simple addition these two expressions may
be combined to give the relationship between
the two most common methods of indicating
an impedance:
R +
jX =IZI (cos
B + j
sin 0)
In the case of impedance, resistance, or reactance, the unit of measurement is the ohm;
hence, the ohm may be thought of as a unit of
opposition to current flow, without reference
to the relative phase angle between the applied voltage and the current which flows.
Further, since both capacitive and inductive
reactance are functions of frequency, impedance will vary with frequency. Figure 12
shows the manner in which IZI will vary with
frequency in an RL series circuit and in an
RC series circuit.
Series RLC Circuits
www.americanradiohistory.com
In a series circuit containing R, L, and C, the im-
HANDBOOK
Impedance
49
pedance is determined as discussed before except that the reactive component in the expressions becomes: (The net reactance-the
difference between XL and Xc.) Hence (XL
Xc) may be substituted for X in the equations.
Thus:
IZI = VR' +(XL
O
= tan
(XL
'(XL
Xc)'
Xc
)
R
A series RLC circuit thus may present an
impedance which is capacitively reactive if
the net reactance is capacitive, inductively
reactive if the net reactance is inductive, or
resistive if the capacitive and inductive reactances are equal.
Addition of
Complex Quantities
The addition of complex
quantities (for example,
impedances in series) is
quite simple if the quantities are in the rectangular form. If they are in the polar form
they only can be added graphically, unless
they are converted to the rectangular form by
the relationships previously given. As an example of the addition of complex quantities
in the rectangular form, the equation for the
addition of impedances is:
( R,
+jX,) +(R, +jX,)= (R,
+ Rs)
+j(X,+X,)
For example if we wish to add the impedances (10 + j50) and (20 j30) we obtain:
(10 + j50) + (20
j30)
= (10 + 20) + j(50 + ( -30)
= 30 + j(50 -30)
It is often necessary in
solving certain types of
circuits to multiply or divide two complex quantities. It is a much simplier mathematical
operation to multiply or divide complex quantities if they are expressed in the polar form.
Hence if they are given in the rectangular
form they should be converted to the polar
form before multiplication or division is begun.
Then the multiplication is accomplished by
multiplying the IZ1 terms together and adding
algebraically the L e terms, as:
(IZ,1Le,)(I4,I Le,)
=IZ,1 Iz,I
L43°)
(1321
L -23 °)
approaches zero.
Division is accomplished by dividing the
denominator into the numerator, and subtracting the angle of the denominator from
that of the numerator, as:
IZ,I Le,
IZ,IL0,-
= 120.321
(L43° + L -23 °)
= 640 L 20°
IZ,I
1z21(Le,
tee,)
For example, suppose that an impedance of
1501 L 67° is to be divided by an impedance
of 1101 L45°. Then:
1501
1101
L67°
L45°
1501
= 151
1101
(L 22 °)
Ohm's Law for
Complex Quantities
The simple form of Ohm's
Law used for d -c circuits
may be stated in a more
general form for application to a -c circuits
involving either complex quantities or simple
resistive elements. The form is:
E
z
(L0,+ L0,)
For example, suppose that the two impedances
1201 L43
and 1321 L -23° are to be multiplied. Then:
( 1201
Figure 12
IMPEDANCE AGAINST FREQUENCY
FOR R L AND R -C CIRCUITS
The impedance of an R -C circuit approaches
infinity os the frequency approaches zero (d.c.),
while the impedance of o series R -L circuit
approaches infinity as the frequency approaches
infinity. The impedance of an R -C circuit approaches the impedance of the series resistor
os the frequency approaches infinity, while the
impedance of o series R -L circuit approaches
the impedance of the resistor as the frequency
)
= 30 + j20
Multiplication and
Division of
Complex Quantities
o
in which, in the general case, I, E, and Z are
complex (vector) quantities. In the simple
case where the impedance is a pure resistance
with an a -c voltage applied, the equation
simplifies to the familiar I = E /R. In any case
the applied voltage may be expressed either
as peak, r.m.s., or average; the resulting
www.americanradiohistory.com
Alternating Current Circuits
50
THE
Since the applied voltage will be the reference
for the currents and voltages within the circuit, we may define it as having a zero phase
angle: E = 100 LO °. Then:
200 n.
oo
_
Figure 13
R -L -C CIRCUIT
=
SERIES
current always will be in the same type of
units as used to define the voltage.
In the more general case vector algebra
must be used to solve the equation. And,
since either division or multiplication is involved, the complex quantities should be expressed in the polar form. As an example,
take the case of the series circuit shown in
figure 13 with 100 volts applied. The impedance of the series circuit can best be obtained
first in the rectangular form, as:
;(l00-.300) = 200
200 + j(100
V200'+(-200)'
= N/40,000 +
=
100 LO °
282 L -45°
40,000
This same current must flow through all three
elements of the circuit, since they are in
series and the current through one must already have passed through the other two.
Hence the voltage drop across the resistor
(whose phase angle of course is 0 °) is:
E
=
The voltage drop across the inductive reactance is:
E = I XL
(0.354 L45 °) (100 L 90 °)
= 35.4 L 135° volts
Similarly, the voltage drop across the capacitive reactance is:
E =
-
E = (0.354 1_45°) (300 /--90 °)
=
-200
X
tan' -=tan'
200
R
=
=
tan' -1
-45 °.
= 282
L -45°
Note that in a series circuit the resulting impedance takes the sign of the largest reactance in the series combination.
Where a slide-rule is being used to make
the computations, the impedance may be found
without any addition or subtraction operations
by finding the angle O first, and then using
the trigonometric equation below for obtaining the impedance. Thus:
O
-tan' -1
=tan'-XR =tan' -200
200
=
-45°
R
Then IZI
=
cos
cos -45°
O
200
IZI
L0°)
70.8 L 45° volts
E = I Xc
80,000
Therefore Z
=1R
E = (0.354 L 45 °) (200
= 282 f2
O=
-0 .354 L0 °- ( -45 °)
0.354 L45° amperes.
-j200
Now, to obtain the current we must convert
this impedance to the polar form.
IZI =
RADIO
0.707
=
0.707
Note that the voltage drop across the capacitive reactance is greater than the supply
voltage. This condition often occurs in a
series RLC circuit, and is explained by the
fact that the drop across the capacitive reactance is cancelled to a lesser or greater extent by the drop across the inductive reactance.
It is often desirable in a problem such as
the above to check the validity of the answer
by adding vectorially the voltage drops across
the components of the series circuit to make
sure that they add up to the supply voltage
or to use the terminology of Kirchhoff's Second
Law, to make sure that the voltage drops
across all elements of the circuit, including
the source taken as negative, is equal to zero.
In the general case of the addition of a
number of voltage vectors in series it is best
to resolve the voltages into their in -phase
and out-of -phase components with respect to
the supply voltage. Then these components
may be added directly. Hence:
-
ER =
=
- 28212
106.2 L -45°
=
=
70.8L45°
70.8 ( cos 45° + j sin 45 °)
70.8 (0.707 + j0.707)
50 + j50
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HANDBOOK
Vector Algebra
51
ao
i
DROP ACROSS RESISTOR
,o
°'-
±leo
45
e
LI NE VOLTAGE =100
PARALLEL CIRCUIT
XL + XL .70.111/-45.
f0
Figure
=
=
=
Ec=
13.
35.4L135°
35.4 ( cos 135° + j sin 135 °)
35.4 ( -0.707 + j0.707)
-25
+ j25
106.2L45°
= 106.2 ( cos -45 °+ j sin -45 °)
= 106.2 (6.707 -j0.707)
= 75
-j75
=(50+ j50)
+(75 -j75)
ER + EL +EC
Figure
+
(-25
ments which go to make up the series circuit
is the same. But the voltage drops across
each of the components are, in general, different from one another. Conversely, in a
parallel RLC or RX circuit the voltage is,
obviously, the same across each of the elements. But the currents through each of the
elements are usually different.
There are many ways of solving a problem
involving paralleled resistance and reactance;
several of these ways will be described. In
general, it may be said that the impedance of
a number of elements in parallel is solved
using the same relations as are used for
solving resistors in parallel, except that complex quantities are employed. The basic re-
lation is:
+
j25)
1
Zrot
the
supply voltage.
It is frequently desirable
to check computations in-
volving complex quantities
by constructing vectors
representing the quantities on the complex
plane. Figure 14 shows such a construction
for the quantities of the problem just completed. Note that the answer to the problem
may be checked by constructing a parallelogram with the voltage drop across the resistor as one side and the net voltage drop
across the capacitor plus the inductor (these
may be added algrebraically as they are 180°
out of phase) as the adjacent side. The vector
sum of these two voltages, which is represented by the diagonal of the parallelogram,
is equal to the supply voltage of 100 volts at
zero phase angle.
Resistance and Reactonce in Parallel
-+
-+ -+
Z,
1
-25+
= (50
75) + j(50 + 25-75)
= 100 +j0
= 100 LO °, which is equal to
Checking by
Construction on the
Complex Plane
15
THE EQUIVALENT SERIES CIRCUIT
Showing a parallel R -C circuit and the equivalent series R -C circuit which represents the
same net impedance os the parallel circuit.
14
Graphical construction of the voltage drops
associated with the serles R -L -C circuit of
figure
EQUIVALENT SERIES CIRCUIT
-43
1
NET DROP ACROSS
,.n
-
.
DROP ACROSS XC =1Oe.2
EL =
T
:_e.i
VOLTAGE DROP ACROSS
X1.= 35.4
laa
series circuit, such
as just discussed, the current through all the ele-
1
1
Z2
Z,
or when only two impedances are involved:
Z, Z2
Z`o` Z t + Z :
As an example, using the two- impedance
relation, take the simple case, illustrated in
figure 15, of a resistance of 6 ohms in parallel with a capacitive reactance of 4 ohms. To
simplify the first step in the computation it is
best to put the impedances in the polar form
for the numerator, since multiplication is involved, and in the rectangular form for the
addition in the denominator.
Zrot
-
(6 L0°) (4
-90°)
6-j4
24
6
L-90°
-j4
Then the denominator is changed to the polar
form for the division operation:
In a
O
=
tañ'
-4
www.americanradiohistory.com
6
=
tan-'
- 0.667 = - 33.7°
THE
Alternating Current Circuits
52
6
IZI =
cos
7.21 ohms
0.832
33.7°
E.
j4 = 7.21 L-33.7°
6
°
Ztat
=
7.21
= 3.33
L
(
Rz
EzEi R1+Rz
L -90°
33.7 °=
cos
= 3.33 [ 0.5548 + j
= 1.85
j
3.33 L -56.3°
56.3° +
j
sin
(- 0.832)1
Through the series of operations in the previous
paragraph we have converted a circuit composed of two impedances in
parallel into an equivalent series circuit composed of impedances in series. An equivalent
series circuit is one which, as far as the terminals are concerned, acts identically to the
original parallel circuit; the current through
the circuit and the power dissipation of the
resistive elements are the same for a given
voltage at the specified frequency.
We can check the equivalent series circuit
of figure 15 with respect to the original circuit by assuming that one volt a.c. (at the
frequency where the capacitive reactance in
the parallel circuit is 4 ohms) is applied to
the terminals of both.
In the parallel circuit the current through
the resistor will be 2/6 ampere (0.166a.) while
the current through the capacitor will be j
ampere (+ j 0.25 a.). The total current will be
the sum of these two currents, or 0.166 +
j 0.25 a. Adding these vectorially we obtain:
W
1
=0.3a.
will
be:
=I2R =0.32x1.85
= 0.9 x 1.85
= 0.166 watts
that the equivalent series circuit
checks exactly with the original parallel circuit.
So we see
Parallel RLC
Circuits
In solving a more complicated
circuit
Ez-E
G+Cz
E2-E1
Lz
LI+Lz
Cz
O
elect to use either of two methods of
solution. These methods are called the admittance method and the assumed - voltage method.
However, the two methods are equivalent
since both use the sum-of-reciprocals equation:
may
1
Ztot
- +- +1
1
1
Z1
Z2
Zs
In the admittance method we use the relation
Y = 1 /Z, where Y = G + jB; Y is called the
admittance, defined above, G is the conductance or R /Z' and B is the susceptance or
X/Z2. Then Ytot = 1 /Ztot = Y1 + Y2 + Y,
In the assumed- voltage method we multiply
both sides of the equation above by E, the
assumed voltage, and add the currents, as:
E
Ztot
-+ -+- ...
E
E
E
Zt
Z,
Z3
= 1zt
+Iz2 +1zt
..
Then the impedance of the parallel combination may be determined from the relation:
Ztot = Eí IZ tot
Voltage dividers for use with
alternating current are quite similar to d-c voltage dividers. However, since capacitors and inductors oppose
the flow of a-c current as well as resistors,
voltage dividers for alternating voltages may
take any of the configurations shown in figDividers
The dissipation in the resistor will be 12/6 =
0.166 watts.
In the case of the equivalent series circuit
the current will be:
And the dissipation in the resistor
xo +502
AC Voltage
III = x/0.1662 + 0.252 = x/0.09 = 0.3 a.
3.33
xCz
Ez=E1
Figure 16
SIMPLE A -C VOLTAGE DIVIDERS
Equivalent Series
Circuit
E
+o
OA
56.3°)
2.77
Ill
,o
Ez
1
Then:
24
I
6
RADIO
made up of more than
two impedances in parallel we
ure 16.
Since the impedances within each divider
are of the same type, the output voltage is in
phase with the input voltage. By using com-
binations of different types of impedances, the
phase angle of the output may be shifted in
relation to the input phase angle at the same
time the amplitude is reduced. Several dividers of this type are shown in figure 17.
Note that the ratio of output voltage to input
voltage is equal to the ratio of the output
impedance to the total divider impedance.
This relationship is true only if negligible
current is drawn by a load on the output terminals.
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HANDBOOK
Xc
E2Ei
Circuits
Resonant
E2
R2+XC2
E
53
XL
R2 +XL2
Figure
©
18
SERIES RESONANT CIRCUIT
If the values of inductance and capacitance
both are fixed, there will be only one resonant
E2E,
XL
Ez
XL-Xc
DO
E,
Ei
Ea
Es - Ei
R2+2
Xc
-
X
R2.2
XL-XCXC
R2+I (L-XC12
COMPLEX
3 -2
Figure 17
VOLTAGE DIVIDERS
A -C
Resonant Circuits
frequency.
If both the inductance and capacitance are
made variable, the circuit may then be changed
or tuned, so that a number of combinations
of inductance and capacitance can resonate at
the same frequency. This can be more easily
understood when one considers that inductive
reactance and capacitive reactance travel in
opposite directions as the frequency is changed.
For example, if the frequency were to remain
constant and the values of inductance and
capacitance were then changed, the following
combinations would have equal reactance:
Frequency is constant at 60 cycles.
L is expressed in henrys.
series circuit such as shown in figure 18
is said to be in resonance when the applied
frequency is such that the capacitive reactance is exactly balanced by the inductive reactance. At this frequency the two reactances
will cancel in their effects, and the impedance
of the circuit will be at a minimum so that
maximum current will flow. In fact, as shown
in figure 19 the net impedance of a series
circuit at resonance is equal to the resistance
which remains in the circuit after the reactances have been cancelled.
A
resistance is always
present in a circuit because it is possessed in some degree by both
the inductor and the capacitor. If the frequency of the alternator E is varied from
nearly zero to some high frequency, there will
be one particular frequency at which the inductive reactance and capacitive reactance
will be equal. This is known as the resonant
frequency, and in a series circuit it is the
frequency at which the circuit current will be
a maximum. Such series resonant circuits are
chiefly used when it is desirable to allow a
certain frequency to pass through the circuit
(low impedance to this frequency), while at
the same time the circuit is made to offer
considerable opposition to currents of other
frequencies.
R
C
is expressed in microfarads (.000001
farad.)
XL
L
.265
2.65
26.5
265.00
2,650.00
1,000
10,000
100,000
1,000,000
Frequency
of Resonance
100
1,000
10.000
100,000
1,000,000
From the formula for resonance, 2rrfL = 1 /2nfC. the resonant frequency is determined:
f=
t Frequency Some
Xc
C
26.5
2.65
.265
.0265
.00265
100
1
2rr
N/
LC
where f = frequency in cycles,
L = inductance in henrys,
C = capacitance in farads.
It is more convenient to express L and C
in smaller units, especially in making radio frequency calculations; f can also be expressed in megacycles or kilocycles. A very
useful group of such formulas is:
f2=
25,330
LC
orL=
25,330
f2C
orC=
25,330
f2L
where f = frequency in megacycles,
L = inductance in microhenrys,
C = capacitance in micromicrofarads.
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54
Alternating Current Circuits
THE
Figure 19
IMPEDANCE OF A
SERIES -RESONANT CIRCUIT
Showing the variation in reactance of the separate elements and In the net impedance of o
series resonant circuit (such as figure 18) with
changing frequency. The vertical line is drawn
at the point of resonance (XL
Xc = 0) in the
-
series circuit.
Impedance of Series
Resonant Circuits
18)
is:
The impedance across
the terminals of a series
resonant circuit (figure
/r3
Z =
Xc)2,
+ (XL
where Z = impedance in ohms,
r = resistance in ohms,
Xc = capacitive reactance in ohms,
XL = inductive reactance in ohms.
From this equation, it can be seen that the
impedance is equal to the vector sum of the
circuit resistance and the difference between
the two reactances. Since at the resonant frequency XL equals Xc. the difference between
them (figure 19) is zero, so that at resonance
the impedance is simply equal to the resistance of the circuit; therefore, because the
resistance of most normal radio- frequency
circuits is of a very low order, the impedance
is also low.
At frequencies higher and lower than the
resonant frequency, the difference between
the reactances will be a definite quantity and
will add with the resistance to make the impedance higher and higher as the circuit is
tuned off the resonant frequency.
If Xc should be greater than XL, then the
term (XL
Xc) will give a negative number.
However, when the difference is squared the
product is always positive. This means that
the smaller reactance is subtracted from the
larger, regardless of whether it be capacitive
or inductive, and the difference squared.
RADIO
FREQUENCY
Figure 20
RESONANCE CURVE
Showing the increase in impedance at resonance for o parallel- resonant circuit, and similarly, the increase in current at resonance for
a series- rsonant circuit. The sharpness of
resonance is determined by the Q of the circuit,
as illustrated by a comparison between A,
B, and C.
Current and Voltage
in Series Resonant
Formulas for calculating
currents and voltages in
a series resonant circuit
are similar to those of
Circuits
Ohm's law.
=-Z
E
I
E =
IZ
The complete equations:
E
I
V' r= +
E =
1
(XL
+
(XL
Xc)2
Xc)'
Inspection of the above formulas will show
the following to apply to series resonant circuits: When the impedance is low, the current
will be high; conversely, when the impedance
is high, the current will be low.
Since it is known that the impedance will
be very low at the resonant frequency, it follows that the current will be a maximum at
this point. If a graph is plotted of the current
against the frequency either side of resonance,
the resultant curve becomes what is known as
a resonance curve. Such a curve is shown in
figure 20, the frequency being plotted against
current in the series resonant circuit.
Several factors will have an effect on the
shape of this resonance curve, of which re-
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Circuit
HANDBOOK
sistance and L -to -C ratio are the important
considerations. The curves B and C in figure
20 show the effect of adding increasing values
of resistance to the circuit. It will be seen
that the peaks become less and less prominent
as the resistance is increased; thus, it can be
said that the selectivity of the circuit is
thereby decreased. Selectivity in this case
can be defined as the ability of a circuit to
discriminate against frequencies adjacent to
the resonant frequency.
Because the a.c. or r -f
voltage across a coil and
capacitor is proportional
to the reactance (for a
given current), the actual voltages across the
coil and across the capacitor may be many
times greater than the terminal voltage of the
circuit. At resonance, the voltage across the
coil (or the capacitor) is Q times the applied
voltage. Since the Q (or merit factor) of a
series circuit can be in the neighborhood of
100 or more, the voltage across the capacitor,
for example, may be high enough to cause
flashover, even though the applied voltage is
of a value considerably below that at which
the capacitor is rated.
Voltage Across Coil
and Capacitor in
Series Circuit
-Sharp-
extremely important
property of a capacitor or
an inductor is its factor of- merit, more generally called its Q. It is this
factor, Q, which primarily determines the
sharpness of resonance of a tuned circuit.
This factor can be expressed as the ratio of
the reactance to the resistance, as follows:
Circuit
Q
An
ness of Resonance
Q-
R
The actual resistance in a wire
or an inductor can be far greater
than the d-c value when the coil is used in a
radio -frequency circuit; this is because the
current does not travel through the entire
cross -section of the conductor, but has a tendency to travel closer and closer to the surface
of the wire as the frequency is increased. This
is known as the skin effect.
The actual current -carrying portion of the
wire is decreased, as a result of the skin
effect, so that the ratio of a -c to d -c resistance of the wire, called the resistance ratio,
is increased. The resistance ratio of wires to
be used at frequencies below about 500 kc.
may be materially reduced through the use of
Utz wire. Litz wire, of the type commonly used
to wind the coils of 455 -kc. i -f transformers,
may consist of 3 to 10 strands of insulated
wire, about No. 40 in size, with the individual
Skin
Effect
Examination of the equation
for determining Q might give
rise to the thought that even
though the resistance of an inductor increases
with frequency, the inductive reactance does
likewise, so that the Q might be a constant.
Actually, however, it works out in practice
that the Q of an inductor will reach a relatively broad maximum at some particular frequency.
Hence, coils normally are designed in such a
manner that the peak in their curve of Q with
frequency will occur at the normal operating
frequency of the coil in the circuit for which
it is designed.
The Q of a capacitor ordinarily is much
higher than that of the best coil. Therefore,
it usually is the merit of the coil that limits
the overall Q of the circuit.
At audio frequencies the core losses in an
iron -core inductor greatly reduce the Q from
the value that would be obtained simply by
dividing the reactance by the resistance. Obviously the core losses also represent circuit
resistance, just as though the loss occurred
in the wire itself.
Variation of Q
with Frequency
circuits, parallel resonance (more correctly termed anti resonance) is more frequently
encountered than series resonance; in fact, it
is the basic foundation of receiver and transmitter circuit operation. A circuit is shown in
figure 21.
Parallel
In radio
Resonance
"Tank" In this circuit, as contrasted with
a circuit for series resonance, L
(inductance) and C (capacitance)
are connected in parallel, yet the combination
can be considered to be in series with the
remainder of the circuit. This combination
of L and C, in conjunction with R, the resistance which is principally included in L, is
sometimes called a tank circuit because it effectively functions as a storage tank when incorporated in vacuum tube circuits.
Contrasted with series resonance, there are
two kinds of current which must be considered
in a parallel resonant circuit: (1) the line current, as read on the indicating meter M (2)
the circulating current which flows within the
parallel L -C -R portion of the circuit. See
figure 21.
At the resonant frequency, the line current
(as read on the meter M,) will drop to a very
low value although the circulating current in
the L -C circuit may be quite large. It is interesting to note that the parallel resonant circuit acts in a distinctly opposite manner to
that of a series resonant circuit, in which the
Circuit
where R = total resistance.
55
strands connected together only at the ends of
the coils.
The
2rrfL
Q
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56
A!ternating Current Circuits
THE
RADIO
plifier circuit, the impedance curve must have
a sharp peak in order for the circuit to be
selective. If the curve is broad- topped in
shape, both the desired signal and the interfering signals at close proximity to resonance
will give nearly equal voltages on the grid of
the tube, and the circuit will then be nonselective; i.e., it will tune broadly.
Figure
21
PARALLEL- RESONANT CIRCUIT
The inductance L and capacitance C comprise
the reactive elements of the parallel -resonant
(anti -resonant) tank circuit, and the resistance
R indicates the sum of the r -f resistance of the
coil and capacitor, plus the resistance coupled
into the circuit from the external load. In most
coses the tuning capacitor has much lower r -f
resistance than the coil and can therefore be
ignored in comparison with the coil resistance
and the coupled -in resistance. The instrument
M1 indicates the "line current" which keeps
the circuit in a state of oscillation
current is the some as the fundamental component
of the plate current of a Class C amplifier which
might be feeding the tank circuit. The instrument M2 indicates the "tank current" which
is equal to the line current multiplied by the
operating Q of the tank circuit.
-this
current is at a maximum and the impedance is
minimum at resonance. It is for this reason
that in a parallel resonant circuit the principal
consideration is one of impedance rather than
current. It is also significant that the impedance curve for parallel circuits is very nearly
identical to that of the current curve for series
resonance. The impedance at resonance is
expressed as:
Z- (2trf
L)2
R
where Z = impedance in ohms,
L = inductance in henrys,
f = frequency in cycles,
R = resistance in ohms.
Or, impedance can be expressed as a function of Q as:
Z= 2nfLQ,
showing that the impedance of a circuit is
directly proportional to its effective Q at
resonance.
The curves illustrated in figure 20 can be
applied to parallel resonance. Reference to the
curve will show that the effect of adding resistance to the circuit will result in both a
broadening out and lowering of the peak of the
curve. Since the voltage of the circuit is
directly proportional to the impedance, and
since it is this voltage that is applied to the
grid of the vacuum tube in a detector or am-
highest
possible voltage can be
developed across a parallel resonant circuit, the impedance of this
circuit must be very high. The impedance will
be greater with conventional coils of limited
Q when the ratio of inductance-to-capacitance
is great, that is, when L is large as compared
with C. When the resistance of the circuit is
In order that the
Effect of L/C Ratio
in Parallel Circuits
very low. XL will equal XC at maximum impedance. There are innumerable ratios of L
and C that will have equal reactance, at a
given resonant frequency, exactly as in the
case in a series resonant circuit.
In practice, where a certain value of inductance is tuned by a variable capacitance
over a fairly wide range in frequency, the
L/C ratio will be small at the lowest frequency end and large at the high -frequency end.
The circuit, therefore, will have unequal gain
and selectivity at the two ends of the band of
frequencies which is being tuned. Increasing
the Q of the circuit (lowering the resistance)
will obviously increase both the selectivity
and gain.
Circulating Tank
The Q of a circuit has
definite bearing on
the
circulating tank
current at resonance. This tank current is
very nearly the value of the line current multiplied by the effective circuit Q. For example:
an r -f line current of 0.050 amperes, with a
circuit Q of 100, will give a circulating tank
current of approximately 5 amperes. From this
it can be seen that both the inductor and the
connecting wires in a circuit with a high Q
must be of very low resistance, particularly in
the case of high power transmitters, if heat
losses are to be held to a minimum.
Because the voltage across the tank at
resonance is determined by the Q, it is possible to develop very high peak voltages
across a high Q tank with but little line current.
Current at Resonance
Effect of Coupling
on Impedance
output circuit, the
Q of the parallel
coupling becomes
(tighter) coupling
www.americanradiohistory.com
a
If a parallel resonant circuit is coupled to another
circuit, such as an antenna
impedance and the effective
circuit is decreased as the
closer. The effect of closer
is the same as though an
Circuit Impedance
HANDBOOK
COUPLING
LOP
O vE N
4E011.14 COUPLING
MEDI U4 0
LOOSE COUPLING
HIGH 0
57
Z
t
Figure
EFFECT OF COUPLING
ON
actual resistance were added in series with
the parallel tank circuit. The resistance thus
coupled into the tank circuit can be considered as being reflected from the output or
load circuit to the driver circuit.
The behavior of coupled circuits depends
largely upon the amount of coupling, as shown
in figure 22. The coupled current in the secondary circuit is small, varying with frequency,
being maximum at the resonant frequency of
the circuit. As the coupling is increased
between the two circuits, the secondary resonance curve becomes broader and the resonant amplitude increases, until the reflected
resistance is equal to the primary resistance.
This point is called the critical coupling
point. With greater coupling, the secondary
resonance curve becomes broader and develops
double resonance humps, which become more
pronounced and farther apart in frequency as
the coupling between the two circuits is
increased.
Tank Circuit
Flywheel Effect
When the plate circuit of a
Class B or Class C operated
tube is connected to a parallel resonant circuit tuned to the same frequency as the exciting voltage for the amplifier, the plate current serves to maintain this
L/C circuit in a state of oscillation.
The plate current is supplied in short pulses
which do not begin to resemble a sine wave,
even though the grid may be excited by a sine wave voltage. These spurts of plate current
are converted into a sine wave in the plate
tank circuit by virtue of the "Q" or "flywheel
effect" of the tank.
If a tank did not have some resistance
losses, it would, when given a "kick" with a
single pulse, continue to oscillate indefinitely.
With a moderate amount of resistance or "fric-
tion" in
the circuit the tank
will still have
22
CIRCUIT IMPEDANCE AND
Q
inertia, and continue to oscillate with decreasing amplitude for a time after being given
a "kick." With such a circuit, almost pure
sine -wave voltage will be developed across
the tank circuit even though power is supplied
to the tank in short pulses or spurts, so long
as the spurts are evenly spaced with respect
to time and have a frequency that is the same
as the resonant frequency of the tank.
Another way to visualize the action of the
tank is to recall that a resonant tank with
moderate Q will discriminate strongly against
harmonics of the resonant frequency. The distorted plate current pulse in a Class C amplifier contains not only the fundamental frequency (that of the grid excitation voltage)
but also higher harmonics. As the tank offers
low impedance to the harmonics and high impedance to the fundamental (being resonant to
a sinethe latter), only the fundamental
appears across the tank circuit
wave voltage
in substantial magnitude.
-
-
Confusion sometimes exists as
to the relationship between the
unloaded and the loaded Q of the
tank circuit in the plate of an r -f power amplifier. In the normal case the loaded Q of the
tank circuit is determined by such factors as
the operating conditions of the amplifier, bandwidth of the signal to be emitted, permissible
level of harmonic radiation, and such factors.
The normal value of loaded Q for an r-f amplifier used for communications service is from
perhaps 6 to 20. The unloaded Q of the tank
circuit determines the efficiency of the output
circuit and is determined by the losses in the
tank coil, its leads and plugs and jacks if any,
and by the losses in the tank capacitor which
ordinarily are very low. The unloaded Q of a
good quality large diameter tank coil in the
high - frequency range may be as high as 500
Loaded and
Unloaded 0
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THE
Alternating Current Circuits
58
to 800, and values greater than 300 are quite
-FUNDAMENTAL SINE WAVE(A
- FUNDAMENTAL
PLUS
3RD HARMONIC(C)
common.
-SQUARE WAVE
Tank Circuit
Since the unloaded Q of a tank
circuit is determined by the
minimum losses in the tank,
while the loaded Q is determined by useful
loading of the tank circuit from the external
load in addition to the internal losses in the
tank circuit, the relationship between the two
Q values determines the operating efficiency
of the tank circuit. Expressed in the form of
an equation, the loaded efficiency of a tank
Efficiency
3RD HARMONIC
Figure
circuit is:
Tank efficiency
=
1
_Qt
x 100
3 -3
23
FUNDAMENTAL PLUS 3RD HARMONIC
unloaded Q of the tank circuit
Qi = loaded Q of the tank circuit
As an example, if the unloaded Q of the
tank circuit for a class C r -f power amplifier
is 400, and the external load is coupled to the
tank circuit by an amount such that the loaded
Q is 20, the tank circuit efficiency will be:
eff. = (1 - 20/400) x 100, or (1 - 0.05) x 100,
or 95 per cent. Hence 5 per cent of the power
output of the Class C amplifier will be lost
as heat in the tank circuit and the remaining
95 per cent will be delivered to the load.
Qu
=
FUNDAMENTAL PLUS 3RD AND
5TH HARMONICS(E)
VI%
HARMONIC
(D)
Figure 24
THIRD HARMONIC WAVE PLUS
FIFTH HARMONIC
FUNDAMENTAL PLUS 3RD. 5TH,
AND 7TH HARMONICS
FUNDAMENTAL PLUS 3RD AND
STM HARMONICS
SQUARE WAVE
(G)
Nonsinusoidal Waves
and Transients
7TM HARMONIC
Pure sine waves,
basic wave shapes.
and complex shape
particularly square
and peaked waves.
(B)
COMPOSITE WAVE-FUNDAMENTAL
PLUS THIRD HARMONIC
Qu
where
RADIO
(F )
discussed previously, are
Waves of many different
are used in electronics,
waves, saw -tooth waves,
Any periodic wave (one that
repeats itself in definite
time intervals) is composed of sine waves of
different frequencies and amplitudes, added
together. The sine wave which has the same
frequency as the complex, periodic wave is
called the fundamental. The frequencies higher
than the fundamental are called harmonics,
and are always a whole number of times higher
than the fundamental. For example, the frequency twice as high as the fundamental is
called the second harmonic.
Wave Composition
The Square Wave
Figure
23 compares a square
wave with a sine wave (A)
of the same frequency. If another sine wave
(B) of smaller amplitude, but three times the
frequency of (A), called the third harmonic, is
added to (A), the resultant wave (C) more
nearly approaches the desired square wave.
Figure 25
RESULTANT WAVE, COMPOSED OF
FUNDAMENTAL, THIRD, FIFTH,
AND SEVENTH HARMONICS
This resultant curve (figure 24) is added to
fifth harmonic curve (D), and the sides of
the resulting curve (E)are steeper than before.
This new curve is shown in figure 25 after a
7th harmonic component has been added to it,
making the sides of the composite wave even
steeper. Addition of more higher odd harmonics
will bring the resultant wave nearer and nearer
a
to the desired square wave shape. The square
wave will be achieved if an infinite number of
odd harmonics are added to the original sine
wave.
www.americanradiohistory.com
Nonsinusoidal
HANDBOOK
FUND.PLUS 2ND ]RD, 4TH,
NAR MENICS
AND
FUND PLUS 210 HARM.
FUNDAMENTAL
2ND HARM.
Waves
59
FUNDAMENTAL PLUS
3RD HARMONIC
FUNDAMENTAL
PLUS 2ND 3RD,
iikkatIPUZTD"'
TM NARMOrMCS
]TM HARMONIC
3RD HARMONIC
-4111111
FVND.
HARMONICS
f YNO.
PLU12Np3RD
ATM\
STM AND STM NARMISN ICS
FUND, PLUS AND AND
PLUS END MARY.
3RD MARYON IC
FUND.
PLUS 2ND, 3RD ATM.
\AND STM MARMONIOS
T
TH HARMONIC
Z
FUND. PLUS 2ND, 3RD,
H XA RM ONI C4
S 2CS AN
A M
&IL"
FUND PLUS 2040, 3R0,4T14,
STM. TH, AND ?TM HARMS.
DL
S:w ÁNDU TMNthreCCaiTa
CATM
7TH HARMONIC
FUNDAMENTAL PLUS 3RD
AND STD HARMONICS
,FUNDAMENTAL PLUS 3RD HARM.
5TH HARMONIC
TOOTH WADE
SA
/
FUND. PLUS 2ND, 3RD4 ATM, STM, ATM,
AND 7TH HARMONICS
Figure 26
COMPOSITION OF A SAWTOOTH WAVE
FUNDAMENTAL PLUS 3RD, STN,
AND 7TH HARMONICS
FUNDAMENTAL PLUS 3RD
ANO STH HARMONIC
the same fashion, a
sawtooth wave is made up
of different sine waves (figure 26). The addition of all harmonics, odd and even, produces
the sawtooth wave form.
The Sawtooth Wave
In
7TH HARMONIC
/
Figure 27 shows the composition of a peaked wave.
Note how the addition of each sucessive harmonic makes the peak of the resultant higher
and the sides steeper.
The Pecked Wave
The three preceeding examples show how a complex
periodic wave is composed of a fundamental
wave and different harmonics. The shape of
the resultant wave depends upon the harmonics
that are added, their relative amplitudes, and
relative phase relationships. In general, the
steeper the sides of the waveform, the more
harmonics it contains.
Figure 27
COMPOSITION OF A PEAKED WAVE
Other Waveforms
If an a -c voltage is substituted for the d-c input voltage in the RC Transient circuits discussed in Chapter 2, the same principles may
be applied in the analysi* of the transient
behavior. An RC coupling circuit is designed
to have a long time constant with respect to
the lowest frequency it must pass. Such a
circuit is shown in figure 28. If a nonsinusoidal voltage is to be passed unchanged
through the coupling circuit, the time constant
AC Transient Circuits
must be long with respect to the period of the
lowest frequency contained in the voltage
wave.
An RC voltage divider that
is designed to distort the input waveform is known as a
differentiator or integrator, depending upon
the locations of the output taps. The output
from a differentiator is taken across the resistance, while the output from an integrator
is taken across the capacitor. Such circuits
will change the shape of any complex a-c
waveform that is impressed upon them. This
distortion is a function of the value of the
time constant of the circuit as compared to
the period of the waveform. Neither a differentiator nor an integrator can change the
RC
Differentiator
and
Integrator
www.americanradiohistory.com
60
Alternating Current Circuits
Cr
0.1 .Uf
100 v.
1000 C.P 5
R
THE
o.M
C'o.1 V
,00v
e.(PEAR)
OUTPUT
VOLTAGE
1000 C.P.S.
R'10R
cc.
RADIO
INTEGRATOR OUTPUT
1T
JJreR'DIPPERENTIATOROUTPUT
50000 USECONDS
R 1lC
PERIOD OP
e' 1000 USECONDS
+100V
Figure 28
R -C COUPLING CIRCUIT WITH
LONG TIME CONSTANT
100 v
et
,-11
INTEGRATOR
OUTPUT
I
E'100v.
(PEAR )
I
I
ófiE[Ñ1TÓR"N
eo
100
V
+25
V.
I
1
10001
-100v
eR
IATOR
DIP
OUTPUT
+30v
PR o
AO
T
OUTPUT Of
ATOR
o
(ec)
T
Figure
30
RC DIFFERENTIATOR AND
INTEGRATOR ACTION ON
Figure 29
R -C DIFFERENTIATOR AND
INTEGRATOR ACTION ON
A SQUARE WAVE
A SINE WAVE
Sawtooth Wave Input
shape of a pure sine wave, they will merely
shift the phase of the wave (figure 29). The
differentiator output is a sine wave leading
the input wave, and the integrator output is a
sine wave which lags the input wave. The sum
of the two outputs at any instant equals the
instantaneous input voltage.
If
a square wave voltage is
impressed on the circuit of
figure 30, a square wave voltage output may
be obtained across the integrating capacitor
if the time constant of the circuit allows the
capacitor to become fully charged. In this
particular case, the capacitor never fully
charges, and as a result the output of the
integrator has a smaller amplitude than the
input. The differentiator output has a maximum
value greater than the input amplitude, since
the voltage left on the capacitor from the
previous half wave will add to the input voltage. Such a circuit, when used as a differentiator, is often called a peaker. Peaks of
twice the input amplitude may be produced.
Square Wave Input
If
a back -to -back saw tooth voltage is applied
to an RC circuit having a time constant one sixth the period of the input voltage, the result is shown in figure 31. The capacitor
voltage will closely follow the input voltage,
if the time constant is short, and the integrator output closely resembles the input. The
amplitude is slightly reduced and there is a
slight phase lag. Since the voltage across the
capacitor is increasing at a constant rate, the
charging and discharging current is constant.
The output voltage of the differentiator, therefore, is constant during each half of the saw tooth input.
voltage waveforms
Various
other than those represented
here may be applied to short
RC circuits for the purpose of producing
across the resistor an output voltage with an
amplitude proportional to the rate of change of
the input signal. The shorter the RC time constant is made with respect to the period of the
input wave, the more nearly the voltage across
Miscellaneous
Inputs
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Transformers
HANDBOOK
e=100
61
INTEGRATOR
OUTPUT (ec)
v.
(PEAR)
1000 C.P.S.
1 DIFFERENTIATOR
JOUTPUT
-
(e0)
+100
OUTPUT WAVEFORM
OF GENERATOR
-100
-ppt{ IO
uJ
e0
ÌA
ERÉITTÓRp
I(eR)
e0
OUTPUT OF
INTEGRATOR
(eE)
Figure 31
DIFFERENTIATOR AND
INTEGRATOR ACTION ON
R -C
A SAWTOOTH WAVE
Figure 32
the capacitor conforms to the input voltage.
Thus, the differentiator output becomes of
particular importance in very short RC circuits. Differentiator outputs for various types
of input waves are shown in figure 32.
The application of a square
wave input signal to audio
equipment, and the observation of the reproduced output signal on
an oscilloscope will provide a quick and accurate check of the overall operation of audio
equipment. Low -frequency and high- frequency
response, as well as transient response can be
examined easily. If the amplifier is deficient
in low- frequency response, the flat top of the
square wave will be canted, as in figure 33.
If the high- frequency response is inferior, the
rise time of the output wave will be retarded
(figure 34). An amplifier with a limited highand low- frequency response will turn the
square wave into the approximation of a saw tooth wave (figure 35).
Square Wave Test
for Audio Equipment
Transformers
are placed in such inductive
coils
When two
relation to each other that the lines of force
3-4
from one cut across the turns of the other
inducing a current, the combination can be
called a transformer. The name is derived from
the fact that energy is transformed from one
winding to another. The inductance in which
Dlfferentlator outputs of short r -c circuits for
various input voltage waveshapes. The output
voltage is proportional to the rote of change
of the input voltage.
the original flux is produced is called the
primary; the inductance which receives the
induced current is called the secondary. In a
radio receiver power transformer, for example,
the coil through which the 110 -volt a.c. passes
is the primary, and the coil from which a higher
or lower voltage than the a -c line potential is
obtained is the secondary.
Transformers can have either air or magnetic cores, depending upon the frequencies at
which they are to be operated. The reader
should thoroughly impress upon his mind the
fact that current can be transferred from one
circuit to another only if the primary current
is changing or alternating. From this it can be
seen that a power transformer cannot possibly
function as such when the primary is supplied
with non -pulsating d.c.
A power transformer usually has a magnetic
core which consists of laminations of iron,
built up into a square or rectangular form,
with a center opening or window. The secondary windings may be several in number, each
perhaps delivering a different voltage. The
secondary voltages will be proportional to the
turns ratio and the primary voltage.
www.americanradiohistory.com
Alternating Current Circuits
62
RADIO
THE
Figure 33
Amplifier deficient In low frequency response will distort square wave applied
to the input circuit, as
shown. A 60 -cycle square wave may he used.
A:
B:
C:
D:
Drop in gain at low frequencies
Lending phase shift at low frequencies
Logging phase shift at low frequencies
Accentuated low frequency gain
Types of
Transformers
Transformers are used in alteroaring- current circuits to transfer power at one voltage and impedance to another circuit at another voltage
and impedance. There are three main classifications of transformers: those made for use
in power-frequency circuits, those made
for
audio -frequency applications, and those made
for radio frequencies.
The Transformation
Ratio
In a perfect transformer all
the magnetic flux lines
produced by the primary
winding link every turn of the secondary winding. For such a transformer, the ratio of the
primary and secondary voltages is exactly
the same as the ratio of the number of turns
in the two windings:
Np
across the secondary
winding
In practice, the transformation ratio of a
transformer is somewhat less than the turns
ratio, since unity coupling does not exist
between the primary and secondary windings.
Ampere Turns (NI)
The current that flows in
the secondary winding as a
result of the induced voltage must produce a
flux which exactly equals the primary flux.
The magnetizing force of a coil is expressed
as the product of the number of turns in the
coil times the current flowing in it:
NpxIp =NsXIs,
or
Np
Ns
=
Is
IP
where Ip = primary current
Ep
Ns
Es
where Np = number of turns in the primary
winding
Ns = number of turns in the secondary
winding
Ep = voltage across the primary winding
Figure
Es = voltage
Is = secondary current
It can be seen from this expression that
when the voltage is stepped up, the current
is stepped down, and vice -versa.
Leakage Reactance
34
Since unity coupling does
not exist in a practical
Figure 35
Output waveshape of amplifier having deficiency
in high- frequency response. Tested with 10 -kc.
square wave.
Output waveshape of amplifier having limited
low -frequency and high- frequency response.
Tested with 1 -kc. square wave.
www.americanradiohistory.com
Electric Filters
HANDBOOK
63
1
STEP -UP
ZL
STEP -OOW N
INPUT
OUTPUT
VOLTAGE
VOLTAGE
Figure 36
IMPEDANCE -MATCHING TRANSFORMER
The reflected impedance Zp varies directly In
the secondary load IL, and
proportion to the square of the
primary-to- secondary turns ratio.
proportion
directly
to
In
transformer, part of the flux passing from the
primary circuit to the secondary circuit follows a magnetic circuit acted upon by the
primary only. The same is true of the secondary flux. These leakage fluxes cause leakage
reactance in the transformer, and tend to
cause the transformer to have poor voltage
regulation. To reduce such leakage reactance,
the primary and secondary windings should
be in close proximity to each other. The more
expensive transformers have interleaved windings to reduce inherent leakage reactance.
Impedance
In the ideal transformer, the
impedance of the secondary
load is reflected back into the
primary winding in the following relationship:
Transformation
Zp = N'Zs , or N = N/Zp/Zs
where Zp = reflected primary impedance
N = turns ratio of transformer
Zs = impedance of secondary load
Thus any specific load connected to the
secondary terminals of the transformer will
be transformed to a different specific value
appearing across the primary terminals of the
transformer. By the proper choice of turns
ratio, any reasonable value of secondary load
impedance may be "reflected" into the primary winding of the transformer to produce the
desired transformer primary impedance. The
phase angle of the primary "reflected" impedance will be the same as the phase angle
of the load impedance. A capacitive secondary load will be presented to the transformer
source as a capacity, a resistive load will
present a resistive "reflection" to the primary
source. Thus the primary source "sees" a
transformer load entirely dependent upon the
secondary load impedance and the turns ratio
of the transformer (figure 36).
The type of transformer in figure
37, when wound with heavy wire
over an iron core, is a common
device in primary power circuits for the purpose of increasing or decreasing the line volt-
Figure 37
THE AUTO -TRANSFORMER
auto- transformer
Schematic diagram of an
showing the method of connecting it to the line
and to the load. When only a small amount of
step up or step down Is required, the auto transformer may be much smaller physically
thon would be a transformer with o separate
Continuously variable
winding.
secondary
auto -transformers (Variar and Powerstat) are
widely used commercially.
age. In effect, it is merely a continuous winding with taps taken at various points along
the winding, the input voltage being applied
to the bottom and also to one tap on the winding. If the output is taken from this same
tap, the voltage ratio will be 1 -to -1; i.e., the
input voltage will be the same as the output
voltage. On the other hand, if the output tap
is moved down toward the common terminal,
there will be a step -down in the turns ratio
with a consequent step-down in voltage. The
initial setting of the middle input tap is chosen
so that the number of turns will have sufficient reactance to keep the no -load primary
current at a reasonably low value.
Electric Filters
3 -5
There are many applications where it is
desirable to pass a d -c component without
passing a superimposed a -c component, or to
ELEMENTARY FILTER SECTIONS
T- NET WONIt
L- SECTIONS
rs
T
Pi - NETWOR
The Auto
Transformer
Figure 38
Complex filters may be mode up from these basic
www.americanradiohistory.com
filter sections.
64
A l t e rn a t
n g
i
C ur r e n
LOW -PASS SHUNT -DERIVE
t
C
i
rc u
i
t
T H E
s
R A D
I
O
HIGH-PASS SERIES -DERI ED FILTER
FILTER
(SERIES -ARM RESONATED
(5J.UNT -ARM RESONATE
CI
2
2CI
2C1
C2
O
<
z
fQ
f2
fq
FREQUENCY
R.
L
FREQUENCY
LOAD RESISTANCE
R.
Ci
1
4M
x C
C2-
K
C2= MCK
LK=
f2 =
LOAD RESISTANCE
CI'
M LE
L2M=
,/I
-()2
CUT -OFF FREQUENCY.
Cx=
777_tF-
fk =FREQUENCY
OF
NIGH ATTENUATION
LK
14M>
-x
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Cs
7- M-I /ßa`2
(
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coverage book.
Filter Operation A filter acts by virtue of its
property of offering very high
impedance to the undesired frequencies, while
offering but little impedance to the desired
frequencies. This will also apply to d.c. with
a superimposed a -c component, as d.c. can
be considered as an alternating current of zero
frequency so far as filter discussion goes.
Basic Filters
Filters are divided into four
classes, descriptive of the fre-
quency bands which they are designed to
transmit: high pass, low pass, band pass and
band elimination. Each of these classes of
filters is made up of elementary filter sections
called L sections which consist of a series
element (ZA) and a parallel element (ZR) as
477fIR
NIGH ATTENUATION
Figure 39
TYPICAL LOW -PASS AND HIGH -PASS FILTERS, ILLUSTRATING
DERIVATIONS
pass all frequencies above or below a certain
frequency while rejecting or attenuating all
others, or to pass only a certain band or bands
of frequencies while attenuating all others.
All of these things can be done by suitable
combinations of inductance, capacitance and
resistance. However, as whole books have
been devoted to nothing but electric filters, it
can be appreciated that it is possible only to
touch upon them superficially in a general
CS
Cur-OFF FREQUENCY. P, =FREQUENCY OF
SHUNT AND SERIES
illustrated in figure 38. A finite number of L
sections may be combined into basic filter
sections, called T networks or pi networks,
also shown in figure 38. Both the T and pi
networks may be divided in two to form halfsections.
Filter Sections
The most common filter section is one in which the two
impedances ZA and Zg are so related that
their arithmetical product is a constant: ZA x
Zg = K2 at all frequencies. This type of filter
section is called a constant-K section.
A section having a sharper cutoff frequency
than a constant -K section, but less attenuation at frequencies far removed from cutoff is
the M- derived section, so called because the
shunt or series element is resonated with a
reactance of the opposite sign. If the complementary reactance is added to the series arm,
the section is said to be shunt derived; if
added to the shunt arm, series derived. Each
impedance of the M- derived section is related
to a corresponding impedance in the constant K section by some factor which is a function
of the constant m. M, in turn, is a function of
the ratio between the cutoff frequency and
the frequency of infinite attenuation, and will
www.americanradiohistory.com
Filter Design
HANDBOOK
TT- SECTION FILTER DESIGN
M=0
CONSTANT K
R' LOAD
RESISTANCE
=CUT-OFF FREQUENCY
2
f.= FREQUENCY OFVERY
HIGH ATTENUATION
°---/-i
C
0
Ln- n R
f2
O
C2
ltf2
R
T
2
f2
RL
fICUT-OFFFREQUENCY
FREQUENCY OF VERY
HIGH ATTENUATION
2L2
LK=4
2L2--
21_2
4
/Vt
I
Cl. S1=
0.6
Lz= la
o.s
R
z
z
ó
0
/¡
='-( f12'oe
)
=
2L2
I
1
2L2 --
}Lt iLt
I
I
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t
I
2L2
0
SAME VALUES
ASM=0.6
1--k
.
1
fm
SAME CURVE AS M
<
\
c
0.6
z
ú
i
1
Lt=3.75L1t=t4M xLK
C/=Cn
L2=LK
H IGH PASS
Cn-
M=0_6
FREQUENCY
CI
LOAD RESISTANCE
o.6
VALUES AS M
--f2
<
.
}SAME
SAME CURVE AS
M
FREQUENCY
CK
o
f
fm
j
j
<
A
4M
Li
TiC2
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T
o o
z
0
0 6
0
t
i
C2=o.6Cn=Cn
z
0
aC
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Ci
yI
t
Lt=0.6Ln=MLK
t-M2
Cz-Cn
.11-(--f-2-,. 2-
f
T
t------0
C2
T
t
Cn -
R.
OTO
Lt=Ln
LOW PASS
M
2
SECTIONSS
LFSEC
NG
-F-YO013y
I
65
f,
.<
r
FREQUENCY
/f
t
FREQUENCY
Figure 40
Through the use of the curves and equations which accompany the diagrams in the illustration above it is
possible to determine the correct values of inductance and capacitance for the usual types of pi- section
filters.
have some value between zero and one. As the
value of m approaches zero, the sharpness of
cutoff increases, but the less will be the
attenuation at several times cutoff frequency.
A value of 0.6 may be used for min most applications. The "notch" frequency is determined
by the resonant frequency of the tuned filter
element. The amount of attenuation obtained
at the "notch" when a derived section is used
is determined by the effective Q of the resonant arm (figure 39).
Filter Assembly
Constant -K sections and derived sections may be cas-
caded to obtain the combined characteristics
of sharp cutoff and good remote frequency
attenuation. Such a filter is known as a composite filter. The amount of attenuation will
depend upon the number of filter sections
used, and the shape of the transmission curve
depends upon the type of filter sections used.
All filters have some insertion loss. This
attenuation is usually uniform to all frequen-
cies within the pass band. The insertion loss
varies with the type of filter, the Q of the
components and the type of termination employed.
Electric wave filters have long
been used in some amateur siations in the audio channel to
reduce the transmission of unwanted high frequencies and hence to reduce the bandwidth
occupied by a radiophone signal. The effectiveness of a properly designed and properly
used filter circuit in reducing QRM and sideband splatter should not be underestimated.
In recent years, high frequency filters have
become commonplace in TVI reduction. High pass type filters are placed before the input
stage of television receivers to reject the
fundamental signal of low frequency transmitters. Low -pass filters are used in the output circuits of low frequency transmitters to
prevent harmonics of the transmitter from
being radiated in the television channels.
Electric Filter
Design
www.americanradiohistory.com
66
Alternating Current Circuits
The chart of figure 40 gives design data
and procedure on the pi- section type of filter.
M- derived sections with an M of 0.6 will be
found to be most satisfactory as the input
section (or half- section) of the usual filter
since the input impedance of such a section
is most constant over the pass band of the
filter section.
Simple filters may use either L, T, or n sections. Since the rr section is the more commonly used type figure 40 gives design data
and characteristics for this type of filter.
A PUSH -PULL 250 -TH AMPLIFIER WITH TVI SHIELD REMOVED
filters in power leads and antenna circuit reduces radiation of TVI- producing harmonics
of typical push -pull amplifier. Shielded enclosure completes harmonic reduction measures.
Use of harmonic
www.americanradiohistory.com
CHAPTE:R FOUR
Vacuum Tube Principles
electron tubes the cathode energy is applied
in the form of heat; electron emission from a
heated cathode is called tbermionic emission.
In another common type of electron tube, the
photoelectric cell, energy in the form of light
is applied to the cathode to cause photoelectric emission.
In the previous chapters we have seen the
manner in which an electric current flows
through a metallic conductor as a result of an
electron drift. This drift, which takes place
when there is a difference in potential between
the ends of the metallic conductor, is in addition to the normal random electron motion
between the molecules of the conductor.
The electron may be considered as a minute
negatively charged particle, having a mass of
9 x 10 -7° gram, and a charge of 1.59 x 10 -19
coulomb. Electrons are always identical,
regardless of the source from which they are
Thermionic Emission
4-1
of electrons from the
cathode of a thermionic electron
tube takes place when the cathode
of the tube is heated to a temperature sufficiently high that the free electrons in the
emitter have sufficient velocity to overcome
the restraining forces at the surface of the
material. These surface forces vary greatly
with different materials. Hence different types
of cathodes must be raised to different temperatures to obtain adequate quantities of electron emission. The several types of emitters
found in common types of transmitting and
receiving tubes will be described in the following paragraphs.
Electron
Emission
obtained.
An electric current can be caused to flow
through other media than a metallic conductor.
One such medium is an ionized solution, such
as the sulfuric acid electrolyte in a storage
battery. This type of current flow is called
electrolytic conduction. Further, it was shown
at about the turn of the century that an electric current can be carried by a stream of free
electrons in an evacuated chamber. The flow
of a current in such a manner is said to take
place by electronic conduction. The study of
electron tubes (also called vacuum tubes, or
valves) is actually the study of the control and
use of electronic currents within an evacuated
or partially evacuated chamber.
Since the current flow in an electron tube
takes place in an evacuated chamber, there
must be located within the enclosure both a
source of electrons and a collector for the
electrons which have been emitted. The electron source is called the cathode, and the
electron collector is usually called the anode.
Some external source of energy must be applied to the cathode in order to impart sufficient velocity to the electrons within the
cathode material to enable them to overcome
the surface forces and thus escape into the
surrounding medium. In the usual types of
Emission
Cathode Types The emitters or cathodes as
used in present -day thermionic electron tubes may be classified into
two groups: the directly- heated or filament
type and the indirectly -heated or heater - cathode
type. Directly- heated emitters may be further
subdivided into three important groups, all
of which are commonly used in modern vacuum
tubes. These classifications are: the puretungsten filament, the thoriated- tungsten
filament, and the oxide-coated filament.
The Pure Tung-
sten Filament
67
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Pure tungsten wire was used
as the filament in nearly all
the earlier transmitting and
68
Vacuum
Tube
THE
Principles
RADIO
Figure
ELECTRON TUBE TYPES
The new General E l e c t r i c ceramic triode (68Y4) is shown alongside a conventional miniature tube (6265) and an octal -based receiving tube (25L6). The ceramic
tube is designed for rugged service and features extremely low lead inductance.
1
receiving tubes. However, the thermionic efficiency of tungsten wire as an emitter (the
number of milliamperes emission per watt of
filament heating power) is quite low, the filaments become fragile after use, their life is
rather short, and they are susceptible to burnout at any time. Pure tungsten filaments must
be run at bright white heat (about 2500° Kelvin). For these reasons, tungsten filaments
have been replaced in all applications where
another type of filament could be used. They
are, however, still universally employed in
large water-cooled tubes and in certain large,
high -power air- cooled triodes where another
filament type would be unsuitable. Tungsten
filaments are the most satisfactory for high power, high -voltage tubes where the emitter
is subjected to positive ion bombardment
caused by the residual gas content of the
tubes. Tungsten is not adversely affected by
such bombardment.
the course of experiments made upon tungsten
emitters, it was found that
filaments made from tungsten having a small
amount of thoria (thorium oxide) as an impurity had much greater emission than those
made from the pure metal. Subsequent development has resulted in the highly efficient carburized thoriated- tungsten filament as used in
virtually all medium -power transmitting tubes
today.
Thoriated-tungsten emitters consist of a
tungsten wire containing from 1% to 2% thoria.
The activation process varies between different manufacturers of vacuum tubes, but
it is essentially as follows: (1) the tube is
evacuated; (2) the filament is burned for a
short period at about 2800° Kelvin to clean
the surface and reduce some of the thoria
within the filament to metallic thorium; (3)
The ThoriatedTungsten Filament
In
the filament is burned for a longer period at
about 2100° Kelvin to form a layer of thorium on the surface of the tungsten; (4) the
temperature is reduced to about 1600° Kelvin
and some pure hydrocarbon gas is admitted
to form a layer of tungsten carbide on the
surface of the tungsten. This layer of tungsten
carbide reduces the rate of thorium evaporation from the surface at the normal operating
temperature of the filament and thus increases
the operating life of the vacuum tube. Thorium evaporation from the surface is a natural
consequence of the operation of the thoriatedtungsten filament. The carburized layer on the
tungsten wire plays another role in acting as
a reducing agent to produce new thorium from
the thoria to replace that lost by evaporation. This new thorium continually diffuses to
the surface during the normal operation of
the filament. The last process, (5), in the
activation of a thoriated tungsten filament consists of re- evacuating the envelope and then
burning or ageing the new filament for a considerable period of time at the normal operating temperature of approximately 1900°K.
One thing to remember about any type of
filament, particularly the thoriated type, is
that the emitter deteriorates practically as
fast when "standing by" (no plate current) as
it does with any normal amount of emission
load. Also, a thoriated filament may be either
temporarily or permanently damaged by a
heavy overload which may strip the surface
layer of thorium from the filament.
Thoriated- tungsten filaments (and only thoriatedtungsten filaments) which
have lost emission as
a result of insufficient filament voltage, a
severe temporary overload, a less severe extended overload, or even normal operation
Reactivating
Thoriated- Tungsten
Filaments
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Types of Emitters
HANDBOOK
Figure
69
2
V -H -F and U -H -F TUBE TYPES
The tube to the left In this photograph is a 955 "acorn" triode. The 6F4 acorn triode is very similar in
appearance to the 955 but has two leads brought out each for the grid and for the plate connection. The
second tubs Is a 446A "lighthouse" triode. The 2C40, 2C43, and 2C44 ore more recent examples of the
same type tube and are
tially the same in external appearance. The third tube from the left is o
2C39 "oilcan" tube. This tube type is essentially the inverse of the lighthouse variety since the cathode
and heater connections come out the small end and the plats is the large finned radiator on the large end.
The use of the finned plate radiator makes the oilcan tube capable of approximately 10 times as much
plate dissipation as the lighthouse type. The tube to the right is the 4X 150A beam tetrads. This tube, a
comparatively recent release, is capable of somewhat greater power output than any of the other tube
types shown, and is rated for full output at 500 Mc. and of reduced output at frequencies greater than
1000 Mc.
may quite frequently be
reactivated to their
original characteristics by a process similar
to that of the original activation. However,
only filaments which have not approached too
close to the end of their useful life may be
successfully reactivated.
The actual process of reactivation is relatively simple. The tube which has gone
"flat" is placed in a socket to which only the
two filament wires have been connected. The
filament is then "flashed" for about 20 to 40
seconds at about 1% times normal rated voltage. The filament will become extremely bright
during this time and, if there is still some
thoria left in the tungsten and if the tube did
not originally fail as a result of an air leak,
some of this thoria will be reduced to metallic
thorium. The filament is then burned at 15 to
25 per cent overvoltage for from 30 minutes to
3 to 4 hours to bring this new thorium to the
surface.
The tube should then be tested to see if it
shows signs of renewed life. If it does, but is
still weak, the burning process should be continued at about 10 to 15 per cent overvoltage
for a few more hours. This should bring it
back almost to normal. If the tube checks still
very low after the first attempt at reactivation,
the complete process can be repeated as a
last effort.
The Oxide.
Coated Filament
The most efficient of all
modern filaments
is the
oxide-coated type which con-
sists of a mixture of barium and strontium
oxides coated upon a nickel alloy wire or
strip. This type of filament operates at a dull red to orange -red temperature (1050' to 1170°
K) at which temperature it will emit large
quantities of electrons. The oxide -coated
filament is somewhat more efficient than the
thoriated- tungsten type in small sizes and it
is considerably less expensive to manufacture.
For this reason all receiving tubes and quite
a number of the low -powered transmitting
tubes use the oxide - coated filament. Another
advantage of the oxide -coated emitter is its
the average tube can be
extremely long life
-
expected to run from 3000 to 5000 hours, and
when loaded very lightly, tubes of this type
have been known to give 50,000 hours of life
before their characteristics changed to any
great extent.
Oxide filaments are unsatisfactory for use
at high continuous plate voltages because: (1)
their activity is seriously impaired by the
high temperature necessary to de -gas the high voltage tubes and, (2) the positive ion bombardment which takes place even in the best
evacuated high -voltage tube causes destruction of the oxide layer on the surface of the
filament.
Oxide -coated emitters have been found capable of emitting an enormously large current
pulse with a high applied voltage for a very
short period of time without damage. This
characteristic has proved to be of great value
www.americanradiohistory.com
70
Vacuum
Tube
Figure 3
CUTAWAY DRAWING OF
A
THE
Principles
6C4 TRIODE
RADIO
age from 2 to 117 volts, although 6.3 is the
most common value. The heater is operated
at quite a high temperature so that the cathode
itself usually may be brought to operating
temperature in a matter of 15 to 30 seconds.
Heat -coupling between the heater and the
cathode is mainly by radiation, although there
is some thermal conduction through the insulating coating on the heater wire, as this
coating is also in contact with the cathode
thimble.
Indirectly heated cathodes are employed in
all a-c operated tubes which are designed to
operate at a low level either for r-f or a -f use.
However, some receiver power tubes use
heater cathodes (6L6, 6V6, 6F6, and 6K6 -GT)
as do some of the low-power transmitter tubes
(802, 807, 815, 3E29, 2E26, 5763, etc.). Heater
cathodes are employed almost exclusively
when a number of tubes are to be operated in
series as in an a.c. -d.c. receiver. A heater
cathode is often called a uni- potential cathode
because there is no voltage drop along its
length as there is in the directly- heated or
filament cathode.
in radar work. For example, the relatively
small cathode in a microwave magnetron may
be called upon to deliver 25 to 50 amperes at
an applied voltage of perhaps 25,000 volts for
a period in the order of one microsecond.
After this large current pulse has been passed,
plate voltage normally will be removed for
1000 microseconds or more so that the cathode
surface may be restored in time for the next
pulse of current. If the cathode were to be
subjected to a continuous current drain of this
magnitude, it would be destroyed in an exceedingly short period of time.
The activation of oxide- coated filaments
also varies with tube manufacturers but consists essentially in heating the wire which has
been coated with a mixture of barium and
strontium carbonates to a temperature of about
1500° Kelvin for a time and then applying a
potential of 100 to 200 volts through a protective resistor to limit the emission current.
This process reduces the carbonates to oxides
thermally, cleans the filament surface of
foreign materials, and activates the cathode
special bombardment cathode is employed in many of
the new high powered television transmitting klystrons(Eimac 3K 20,000
LA). The cathode takes the form of a tantalum
diode, heated to operating temperature by the
bombardment of electrons from a directly
heated filament. The cathode operates at a
positive potential of 2000 volts with respect
to the filament, and a d-c bombardment current of 0.66 amperes flows between filament
and cathode. The filament is designed to
operate under space -charge limited conditions.
Cathode temperature is varied by changing the
bombardment potential between the filament
and the cathode.
The heater type cathode was developed as a result of the requirement for a type of emitter
which could be operated from alternating current and yet would not introduce a-c ripple
modulation even when used in low -level stages.
It consists essentially of a small nickel -alloy
cylinder with a coating of strontium and barium oxides on its surface similar to the coating used on the oxide- coated filament. Inside
the cylinder is an insulated heater element
consisting usually of a double spiral of tungsten wire. The heater may operate on any volt-
The emission of electrons from
a heated cathode is quite similar to the evaporation of molecules from the surface of a liquid. The molecules which leave the surface are those having
sufficient kinetic (heat) energy to overcome
the forces at the surface of the liquid. As the
temperature of the liquid is raised, the average velocity of the molecules is increased,
and a greater number of molecules will acquire
sufficient energy to be evaporated. The evaporation of electrons from the surface of a thermionic emitter is similarly a function of average electron velocity, and hence is a function
of the temperature of the emitter.
Electron emission per unit area of emitting
surface is a function of the temperature T
in degrees Kelvin, the work function of the
emitting surface b (which is a measure of the
surface.
Reactivation of oxide- coated filaments is
not possible since there is always more than
sufficient reduction of the oxides and diffusion
of the metals to the surface of the filament to
meet the emission needs of the cathode.
The Heater
Cathode
The Bombardment
Cathode
The Emission
Equation
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A
Thermionic Emission
HANDBOOK
71
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CUT -AWAY DRAWING OF A 6CB6 PENTODE
surface forces of the material and hence of
the energy required of the electron before it
may escape), and of the constant A which
also varies with the emitting surface. The relationship between emission current in amperes per square centimeter, 1, and the above
quantities can be expressed as:
= AT'c"b'T
Secondary The bombarding of most metals
and a few insulators by electrons
Emission
will result in the emission of other
1
electrons
by a
process called secondary emis-
The secondary electrons are literally
knocked from the surface layers of the bombarded material by the primary electrons which
strike the material. The number of secondary
electrons emitted per primary electron varies
from a very small percentage to as high as
5 to 10 secondary electrons per primary.
The phenomena of secondary emission is
undesirable for most thermionic electron tubes.
However, the process is used to advantage in
certain types of electron tubes such as the
image orthicon (TV camera tube) and the
electron -multiplier type of photo -electric cell.
In types of electron tubes which make use of
secondary emission, such as the type 931
photo cell, the secondary- electron -emitting
surfaces are specially treated to provide a
high ratio of secondary to primary electrons.
Thus a high degree of current amplification in
the electron -multiplier section of the tube is
sion.
obtained.
As a cathode is heated so that
it begins to emit, those electrons which have been discharged into the surrounding space form a
negatively charged cloud in the immediate
vicinity of the cathode. This cloud of electrons
around the cathode is called the space charge.
The electrons comprising the charge are conThe Space
Charge Effect
tinuously
changing,
since those electrons
making up the original charge fall back into
20
10
30
ao
so
D.C. PLATE VOLTS
Figure
5
AVERAGE PLATE CHARACTERISTICS
OF A POWER DIODE
the cathode and are replaced by others emitted
by it.
4 -2
The Diode
If a cathode capable of being heated either
indirectly or directly is placed in an evacuated
envelope along with a plate, such a two element vacuum tube is called a diode. The
diode is the simplest of all vacuum tubes and
is the fundamental type from which all the
others are derived.
the cathode within a
diode is heated, it will be
found that a few of the electrons leaving the cathode will leave with sufficient velocity to reach the plate. If the plate
is electrically connected back to the cathode,
the electrons which have had sufficient veloc=
ity to arrive at the plate will flow back to the
cathode through the external circuit. This
small amount of initial plate current is an
effect found in all two -element vacuum tubes.
If a battery or other source of d -c voltage
is placed in the external circuit between the
plate and cathode so that it places a positive
potential on the plate, the flow of current from
the cathode to plate will be increased. This is
due to the strong attraction offered by the positively charged plate for any negatively charged
particles (figure 5).
Characteristics
of the Diode
When
At moderate values of
plate voltage the current flow from cathode
to anode is limited by the space charge of
electrons around the cathode. Increased values
Space- Charge Limited
Current
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Vacuum
72
Principles
Tube
THE
RADIO
DE COATED
ED TUNGSTEN
TUNGSTEN FILAMENT
POINT OF MAXIMUM SPACE CHARGE -LIMITED EMISSION
Figure
ACTION
PLATE VOLTAGE
Figure
+
6
MAXIMUM SPACE -CHARGE -LIMITED
EMISSION FOR DIFFERENT
TYPES OF EMITTERS
of plate voltage will tend to neutralize a
greater portion of the cathode space charge
and hence will cause a greater current to flow.
Under these conditions, with plate current
limited by the cathode space charge, the plate
current is not linear with plate voltage. In
fact it may be stated in general that the plate current flow in electron tubes does not obey
Ohm's Law. Rather, plate current increases as
the three -halves power of the plate voltage.
The relationship between plate voltage, E,
and plate current, 1, can be expressed as:
/
=K
F3!2
where K is a constant determined by the
geometry of the element structure within the
electron tube.
As plate voltage is raised to
the potential where the cathode space charge is neutralized, all the electrons that the cathode is capable of emitting are being attracted to the
plate. The electron tube is said then to have
reached saturation plate current. Further increase in plate voltage will cause only a
relatively small increase in plate current. The
initial point of plate current saturation is
sometimes called the point of Maximum Space Charge- Limited Emission (MSCLE).
The degree of flattening in the plate -voltage
plate- current curve after the MSCLE point will
vary with different types of cathodes. This effect is shown in figure 6. The flattening is
quite sharp with a pure tungsten emitter. With
thoriated tungsten the flattening is smoothed
somewhat, while with an oxide- coated cathode
the flattening is quite gradual. The gradual
saturation in emission with an oxide- coated
emitter is generally considered to result from
Plate Current
Saturation
7
OF
THE GRID IN A TRIODE
(A) shows the triode tube with cutoff bias on
the grid. Note that all the electrons emitted
by the cathode remain inside the grid mesh.
(B) shows the same tube with an intermediate
value of bias on the grid. Note the medium
value of plate current and the fact that there
is a reserve of electrons remaining within the
grid mesh. (C) shows the operation with a
relatively small amount of bias which with
certain tube types will allow substantially all
the electrons emitted by the cathode to reach
the plate. Emission is said to be saturated in
this case. In a majority of tube types a high
value of positive grid voltage is required before plate - current saturation takes place.
a lowering of the surface work function by the
field at the cathode resulting from the plate
potential.
Electron Energy
Dissipation
The current flowing in the
plate- cathode space of a conducting electron tube represents the energy required to accelerate electrons from the zero potential of the cathode
space charge to the potential of the anode.
Then, when these accelerated electrons strike
the anode, the energy associated with their
velocity is immediately released to the anode
structure. In normal electron tubes this energy
release appears as heating of the plate or
anode structure.
4-3
The Triode
If an element consisting of a mesh or spiral
of wire is inserted concentric with the plate
and between the plate and the cathode, such
an element will be able to control by electrostatic action the cathode -to -plate current of
the tube. The new element is called a grid, and
a vacuum tube containing a cathode, grid, and
plate is commonly called a triode.
Action of
If this new element through which
the electrons must pass in their
course from cathode to plate is made
negative with respect to the cathode, the negathe Grid
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HANDBOOK
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Current Flow
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500
PLATE VOLTS (EP)
Figure 8
NEGATIVE -GRID CHARACTERISTICS(Ip
VS. Ep CURVES) OF A
TYPICAL
TRIODE
Average plate characteristics of this type
are most commonly used in determining the
Class A operating characteristics of a
triode amplifier stage.
cive charge on this grid will effectively repel
the negatively charged electrons (like charges
repel; unlike charges attract) back into the
space charge surrounding the cathode Hence,
the number of electrons which are able to pass
through the grid mesh and reach the plate will
be reduced, and the plate current will be reduced accordingly. If the charge on the grid
is made sufficiently negative, all the electrons
leaving the cathode will be repelled back to
it and the plate current will be reduced to zero.
Any d-c voltage placed upon a grid is called
a bias (especially so when speaking of a control grid). The smallest negative voltage which
will cause cutoff of plate current at a particular plate voltage is called the value of cutoff
bias (figure 7).
Amplification The amount of plate current in a
Factor
triode is a result of the net field
at the cathode from interaction
between the field caused by the grid bias and
that caused by the plate voltage. Hence, both
grid bias and plate voltage affect the plate
current. In all normal tubes a small change in
grid bias has a considerably greater effect
than a similar change in plate voltage. The
ratio between the change in grid bias and the
change in plate current which will cause the
same small change in plate current is called
the amplification /actor or
of the electron
tube. Expressed as an equation:
AE
- AE,
73
with i, constant (A represents a small increment).
The µ can be determined experimentally by
making a small change in grid bias, thus
slightly changing the plate current. The plate
current is then returned to the original value
by making a change in the plate voltage. The
ratio of the change in plate voltage to the
change in grid voltage is the µ of the tube
under the operating conditions chosen for the
16
s!
Characteristics
Triode
In a diode it was shown that
the electrostatic field at the
cathode was proportional to
the plate potential, Ep, and that the total
cathode current was proportional to the three halves power of the plate voltage. Similiarly,
in a triode it can be shown that the field at
the cathode space charge is proportional to
the equivalent voltage (Eg + Ep /ft), where
the amplification factor, µ, actually represents
the relative effectiveness of grid potential and
plate potential in producing a field at the
cathode.
It would then be expected that the cathode
current in a triode would be proportional to
the three -halves power of (E5 + Ep /µ). The
cathode current of a triode can be represented
with fair accuracy by the expression:
Cathode current
= K
(E5
E
3/2
+-=-)
)
where K is a constant determined by element
geometry within the triode.
Plate Resistance The plate resistance of a
vacuum tube is the ratio of a
change in plate voltage to the change in plate
current which the change in plate voltage
produces. To be accurate, the changes should
be very small with respect to the operating
values. Expressed as an equation:
Rp =
AE P
p
E,=
constant, A = small
increment
The plate resistance can also be determined
by the experiment mentioned above. By noting
the change in plate current as it occurs when
the plate voltage is changed (grid voltage
held constant), and by dividing the latter by
the former, the plate resistance can be determined. Plate resistance is expressed in Ohms.
The mutual conductance,
also referred to as trans conductance, is the ratio of a change in the
plate current to the change in grid voltage
which brought about the plate current change,
the plate voltage being held constant. Expressed as an equation:
Transconductance
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Tube
Vacuum
74
Principles
RADIO
THE
430
400
330
300
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GRID VOLTAGE (E9)
70
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100
characteristics of this type are most
commonly used in determining the pulse -signal
operating characteristics of a triode amplifier
stage. Note the large emission capability of
the oxidecoated heater cathode in tubes of the
type of the 6J5.
9
Al
constant,
small
increment
The transconductance is also numerically
equal to the amplification factor divided by
the plate resistance. Gm =
E
=
A =
AEg
Transconductance is most commonly expressed in microreciprocal -ohms or micro mhos. However, since transconductance expresses change in plate current as a function
of a change in grid voltage, a tube is often
said to have a transconductance of so many
milliamperes- per-volt. If the transconductance
in milliamperes -per-volt is multiplied by 1000
it will then be expressed in micromhos. Thus
the transconductance of a 6A3 could be called
either 5.25 ma.!volt or 5250 micromhos.
0
1-5
(Es)
Figure 10
This type of graphical
for Class C amplifier
operating characteristic
is a straight line when
Plate
1
-10
-5
GRID VOLTS
CONSTANT CURRENT (Ep VS. E9)
CHARACTERISTICS OF A
TYPICAL TRIODE TUBE
Figure 9
POSITIVE -GRID CHARACTERISTICS
(Ip VS. E9) OF A TYPICAL TRIODE
Gm =
-11
representation is used
calculations since the
of a Class C amplifier
drawn upon a constant
current- graph.
passing through the plate circuit of the tube
for various values of plate -load resistance and
plate - supply voltage. Figure 11 illustrates a
triode tube with a resistive plate load, and a
supply voltage of 300 volts. The voltage at
the plate of the tube (ep) may be expressed
as:
ep = Ep
-(i
x RL)
where Ep is the plate supply voltage, ip is the
plate current, and RL is the load resistance in
ohms.
Assuming various values of ip flowing in
the circuit, controlled by the internal resistance of the tube, (a function of the grid bias)
values of plate voltage may be plotted as
shown for each value of plate current (ir). The
line connecting these points is called the
load line for the particular value of plate -load
resistance used. The slope of the load line is
equal to the ratio of the lengths of the vertical
and horizontal projections of any segment of
the load line. For this example it is:
The operating character istics of a triode tube
may be summarized in
three sets of curves: The Ip vs. Ep curve
(figure 8), the Ip vs. Eg curve (figure 9) and
the Ep vs. E curve (figure 10). The plate
resistance (Rj of the tube may be observed
from the Ip vs. Ep curve, the transconductance
(Gm) may be observed from the Ip vs. Eg curve,
and the amplification factor (pt) may be determined from the Ep vs. Eg curve.
The slope of the load line is equal to
-1/11L. At point A on the load line, the voltage across the tube is zero. This would be
true for a perfect tube with zero internal voltage drop, or if the tube is short -circuited from
cathode to plate. Point B on the load line
load line is a graphical
representation of the voltage
on the plate of a vacuum tube, and the current
corresponds to the cutoff point of the tube,
where no plate current is flowing. The operating range of the tube lies between these
two extremes. For additional information re-
Characteristic Curves
of a Triode Tube
The Load Line
A
Slope
=
-
www.americanradiohistory.com
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100
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200
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10,000
Triode Load Line
HANDBOOK
IP(YA)i En
S
10
IS
20
25
30
EP=300v
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300
250
200
I
75
RL=en
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100
50
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Figure 12
TRIODE TUBE CONNECTED FOR DETERMINATION OF PLATE CIRCUIT LOAD
LINE, AND OPERATING PARAMETERS
OF THE CIRCUIT
300
EP
Figure
11
voltage drop across the plate load resistor,
RL. The plate voltage on the tube is therefore
300 volts. If, on the other hand, the tube is
considered to be a short circuit, maximum
possible plate current flows and the full 300
volt drop appears across RL. The plate voltage is zero, and the plate current is 300/8,000,
or 37.5 milliamperes. These two extreme conditions define the load line on the I, vs. Ep
characteristic curve, figure 13.
For this application the grid of the tube is
returned to a steady biasing voltage of -4
volts. The steady or quiescent operation of the
tube is determined by the intersection of the
load line with the -4 volt curve at point Q.
By projection from point Q through the plate
The static load line for a typical triode
tube with a plate load resistance of 10,000
ohms.
garding dynamic load lines, the reader is
referred to the Radiotron Designer's Handbook,
4th edition, distributed by Radio Corporation
of America.
Application of Tube
Characteristics
As an example of the ap-
plication of tube characteristics, the constants
of the triode amplifier circuit shown in figure
12
volts, and the plate load is 8,000 ohms.
If the tube is considered to be an open circuit
no plate current will flow, and there is no
may be considered. The plate supply is
40
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APPLICATION OF Ip VS. Ep
CHARACTERISTICS OF
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s
300
400
PLATE VOLTS (E.)
N
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VOLT Pt.AT[ SWING
`
Vacuum
76
EG
-4r
Tube
\
Principles
. D.C.
THE
RADIO
BIAS LEVEL (EC)
TFigure 15
SCHEMATIC REPRESENTATION
+ 18.23
OF
STEADY STATE //
PLATE CURRENT\i)1
INTERELECTRODE
CAPACITANCE
+is
comes apparent. A voltage variation of 8 volts
(peak -to -peak) on the grid produces a variation
of 84 volts at the plate.
T-
Polarity Inversion
STEADY STATE (EP)
PLATE VOLTAGE
EP
Figure
14
POLARITY REVERSAL BETWEEN GRID
AND PLATE VOLTAGES
current axis it is found that the value of plate
current with no signal applied to the grid is
12.75 milliamperes. By projection from point
Q through the plate voltage axis it is found
that the quiescent plate voltage is 198 volts.
This leaves a drop of 102 volts across RL
which is borne out by the relation 0.01275 x
8,000 = 102 volts.
An alternating voltage of 4 volts maximum
swing about the normal bias value of -4 volts
is applied now to the grid of the triode amplifier. This signal swings the grid in a positive
direction to 0 volts, and in a negative direction
to -8 volts, and establishes the operating
region of the tube along the load line between
points A and B. Thus the maxima and minima
of the plate voltage and plate current are
established. By projection from points A and
B through the plate current axis the maximum
instantaneous plate current is found to be
18.25 milliamperes and the minimum is 7.5
milliamperes. By projections from points A and
B through the plate voltage axis the minimum
instantaneous plate voltage swing is found to
be 154 volts and the maximum is 240 volts.
By this graphical application of the IP vs.
Ep characteristic of the 6SN7 triode the operation of the circuit illustrated in figure 12 be-
signal voltage applied to the grid has its
maximum positive instantaneous value the
plate current is also maximum. Reference to
figure 12 shows that this maximum plate current flows through the plate load resistor RL,
producing a maximum voltage drop across it.
The lower end of RL is connected to the plate
supply, and is therefore held at a constant
potential of 300 volts. With maximum voltage
drop across the load resistor, the upper end of
RL is at a minimum instantaneous voltage.
The plate of the tube is connected to this end
of RL and is therefore at the same minimum
instantaneous potential.
This polarity reversal between instantaneous
grid and plate voltages is further clarified by
a consideration of Kirchhoff's law as it applies to series resistance. The sum of the IR
drops around the plate circuit must at all times
equal the supply voltage of 300 volts. Thus
when the instantaneous voltage drop across
RL is maximum, the voltage drop across the
tube is minimum, and their sum must equal
300 volts. The variations of grid voltage,plate
current and plate voltage about their steady
state values is illustrated in figure 14.
When the
Capacitance always exists between any two pieces of metal
separated by a dielectric. The
exact amount of capacitance depends upon the
size of the metal pieces, the dielectric between them, and the type of dielectric. The
electrodes of a vacuum tube have a similar
characteristic known as the interelectrode
capacitance, illustrated in figure 15. These
direct capacities in a triode are: grid-tocathode capacitance, grid -to-plate capacitance,
and plate -to- cathode capacitance. The interelectrode capacitance, though very small, has
a coupling effect, and often can cause unbalance in a particular circuit. At very high
Interelectrode
Capacitance
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Tetrodes
HANDBOOK
Pentodes
and
TYPE 24-A
77
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6
TYPE 6SK7
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200
300
S00
100
VOLTS (Eel
Figure
TYPICAL
16
200
300
VOLTS
(E)
Figure
TETRODE
CHARACTERISTIC CURVES
Ip VS. Ep
frequencies (v-h -f), interelectrode capacities
become very objectionable and prevent the use
of conventional tubes at these frequencies.
Special v -h -f tubes must be used which are
characterized by very small electrodes and
close internal spacing of the elements of the
tube.
400
500
17
TYPICAL IP VS. EP PENTODE
CHARACTERISTIC CURVES
the electrons pass through it and on to the
plate. Due also to the screen, the plate current is largely independent of plate voltage,
thus making for high amplification. When the
screen voltage is held at a constant value, it
is possible to make large changes in plate
voltage without appreciably affecting the plate
current, (figure 16).
4 -4
Tetrode or Screen Grid Tubes
Many desirable characteristics can be obtained in a vacuum tube by the use of more
than one grid. The most common multi -element
tube is the tetrode (four electrodes). Other
tubes containing as many as eight electrodes
are available for special applications.
The quest for a simple and easily
usable method of eliminating the
effects of the grid-to -plate capacitance of the
triode led to the development of the screen grid tube or tetrode. When another grid is
added between the grid and plate of a vacuum
tube the tube is called a tetrode, and because
the new grid is called a screen, as a result of
its screening or shielding action, the tube is
often called a screen -grid tube. The interposed screen grid acts as an electrostatic
shield between the grid and plate, with the
consequence that the grid-to -plate capacitance
is reduced. Although the screen grid is maintained at a positive voltage with respect to
the cathode of the tube, it is maintained at
ground potential with respect to r.f. by means
of a by-pass capacitor of very low reactance
at the frequency of operation.
In addition to the shielding effect, the
screen grid serves another very useful purpose.
Since the screen is maintained at a positive
potential, it serves to increase or accelerate
the flow of electrons to the plate. There being
large openings in the screen mesh, most of
The Tetrode
When the electrons from the cathode approach the plate with sufficient velocity, they
dislodge electrons upon striking the plate.
This effect of bombarding the plate with high
velocity electrons, with the consequent dislodgement of other electrons from the plate,
gives rise to the condition of secondary emission which has been discussed in a previous
paragraph. This effect can cause no particular
difficulty in a triode because the secondary
electrons so emitted are eventually attracted
back to the plate. In the screen -grid tube, however, the screen is close to the plate and is
maintained at a positive potential. Thus, the
screen will attract these electrons which have
been knocked from the plate, particularly when
the plate voltage falls to a lower value than
the screen voltage, with the result that the
plate current is lowered and the amplification
is decreased.
In the application of tetrodes, it is necessary to operate the plate at a high voltage in
relation to the screen in order to overcome
these effects of secondary emission.
The undesirable effects of secondary emission from the plate
can be greatly reduced if yet another element
is added between the screen and plate. This
additional element Is called a suppressor, and
tubes in which it is used are called pentodes.
The suppressor grid is sometimes connected
to the cathode within the tube; sometimes it is
brought out to a connecting pin on the tube
base, but in any case it is established negaThe Pentode
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Vacuum
78
Tube
Principles
THE
RADIO
GRiD
-
C
HODE
.L-
Cÿ
REMOTE CUT -OFF
GRID
SHARP CUT -OFF
GRID
-
Figure 18
REMOTE CUTOFF GRID STRUCTURE
tive with respect to the minimum plate voltage. The secondary electrons that would travel
to the screen if there were no suppressor are
diverted back to the plate. The plate current
is, therefore, not reduced and the amplification possibilities are increased (figure 17).
Pentodes for audio applications are designed so that the suppressor increases the
limits to which the plate voltage may swing;
therefore the consequent power output and
gain can be very great. Pentodes for radio frequency service function in such a manner
that the suppressor allows high voltage gain,
at the same time permitting fairly high gain
at low plate voltage. This holds true even if
the plate voltage is the same or slightly lower
than the screen voltage.
Remote cutoff tubes (variable
are screen grid tubes in
which the control grid structure has been physically modified so as to
cause the plate current of the tube to drop off
gradually, rather than to have a well defined
cutoff point (figure 18). A non -uniform control
grid structure is used, so that the amplification factor is different for different parts of the
Remote Cutoff
mu)
Tubes
control grid.
Remote cutoff tubes are used in circuits
where it is desired to control the amplification
by varying the control grid bias. The characteristic curve of an ordinary screen grid tube
has considerable curvature near the plate current cutoff point, while the curve of a remote
cutoff tube is much more linear (figure 19).
The remote cutoff tube minimizes cross-
talk
interference that would otherwise be
produced. Examples of remote cutoff tubes
are: 6BD6, 6K7, 6SG7 and 6SK7.
beam power tube makes use
of another method for suppressing
secondary emission. In this tube
there are four electrodes: a cathode, a grid, a
screen, and a plate, so spaced and placed that
secondary emission from the plate is suppressed without actual power loss. Because
Beam Power
Tubes
A
GRID VOLTS
Figure 19
A REMOTE CUTOFF
GRID STRUCTURE
ACTION OF
of the manner in which the electrodes are
spaced, the electrons which travel to the
plate are slowed down when the plate voltage
is low, almost to zero velocity in a certain
region between screen and plate. For this
reason the electrons form a stationary cloud,
or space charge. The effect of this space
charge is to repel secondary electrons emitted
from the plate and thus cause them to return
to the plate. In this way, secondary emission
is suppressed.
Another feature of the beam power tube is
the low current drawn by the screen. The
screen and the grid are spiral wires wound so
that each turn in the screen is shaded from
the cathode by a grid turn. This alignment of
the screen and the grid causes the electrons
to travel in sheets between the turns of the
screen so that very few of them strike the
screen itself. This formation of the electron
stream into sheets or beams increases the
charge density in the screen -plate region and
assists in the creation of the space charge in
this region.
Because of the effective suppressor action
provided by the space charge, and because of
the low current drawn by the screen, the beam
power tube has the advantages of high power
output, high power -sensitivity, and high efficiency. The 6L6 is such a beam power tube,
designed for use in the power amplifier stages
of receivers and spec -h amplifiers or modulators. Larger tubes employing the beam -power
principle are being made by various manufacturers for use in the radio -frequency stages
of transmitters. These tubes feature extremely
high power- sensitivity (a very small amount
of driving power is required for a large output), good plate efficiency, and low grid -toplate capacitance. Examples of these tubes
are 813, 4 -250A, 4X150A, etc.
The grid- screen mu factor (Ass)
is analogous to the amplification
factor in a triode, except that
the screen of a pentode or tetrode is subGrid -Screen
Mu
Factor
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HANDBOOK
Mixer and Converter
stituted for the plate of a triode. µ5g denotes
the ratio of a change in grid voltage to a
change in screen voltage, each of which will
produce the same change in screen current.
Expressed as an equation:
AEss
flag =
AEs
Ise = constant, A = small
increment
The grid- screen mu factor is important in
determining the operating bias of a tetrode
or pentode tube. The relationship between control -grid potential and screen potential determines the plate current of the tube as well as
the screen current since the plate current is
essentially independent of the plate voltage
in tubes of this type. In other words, when
the tube is operated at cutoff bias as determined by the screen voltage and the grid screen mu factor (determined in the same way
as with a triode, by dividing the operating
voltage by the mu factor) the plate current
will be substantially at cutoff, as will be the
screen current. The grid- screen mu factor is
numerically equal to the amplification factor
of the same tetrode or pentode tube when
it is triode connected.
The following equation is the
expression for total cathode cur rent in a triode tube. The expression for the total cathode
current of a tetrode and a pentode tube is the
same, except that the screen -grid voltage and
the grid- screen it-factor are used in place of
the plate voltage and it of the triode.
Current Flow
in Tetrodes
and Pentodes
Cathode current = K
/
1
E
Es +
3/2
$g )
Ilse
Cathode current, of course, is the sum of the
screen and plate current, plus control grid current in the event that the control grid is positive with respect to the cathode. It will be
noted that total cathode current is independent
of plate voltage in a tetrode or pentode. Also,
in the usual tetrode or pentode the plate current is substantially independent of plate
voltage over the usual operating range- which
means simply that the effective plate resistance of such tubes is relatively high. However, when the plate voltage falls below the
normal operating range, the plate current
falls sharply, while the screen current rises to
such a value that the total cathode current
remains substantially constant. Hence, the
screen grid in a tetrode or pentode will almost
invariably be damaged by excessive dissipation if the plate voltage is removed while the
screen voltage is still being applied from a
low -impedance source.
Tubes
79
The current equations show how
the total cathode current in
triodes, tetrodes, and pentodes
is a function of the potentials applied to the
various electrodes. If only one electrode is
positive with respect to the cathode (such as
would be the case in a triode acting as a
class A amplifier) all the cathode current goes
to the plate. But when both screen and plate
are positive in a tetrode or pentode, the cathode current divides between the two elements.
Hence the screen current is taken from the
total cathode current, while the balance goes
to the plate. Further, if the control grid in a
tetrode or pentode is operated at a positive
potential the total cathode current is divided
between all three elements which have a positive potential. In a tube which is receiving a
large excitation voltage, it may be said that
the control grid robs electrons from the output
electrode during the period that the grid is
positive, making it always necessary to limit
the peak -positive excursion of the control
grid.
The Effect of
Grid Current
In general it may be stated
that the amplification factor
of tetrode and pentode tubes
is a coefficient which is not
of much use to the designer. In fact the amplification factor is seldom given on the design
data sheets of such tubes. Its value is usually
very high, due to the relatively high plate
resistance of such tubes, but bears little
relationship to the stage gain which actually
will be obtained with such tubes.
On the other hand, the grid-plate transconductance is the most important coefficient of
pentode and tetrode tubes. Gain per stage can
be computed directly when the Gm is known.
The grid -plate transconductance of a tetrode
or pentode tube can be calculated through use
of the 'expression:
Alp
Coefficients of
Tetrodes and
Pentodes
Gm
=
AE e
with E5s and Es constant.
The plate resistance of such tubes is of
less importance than in the case of triodes,
though it is often of value in determining the
amount of damping a tube will exert upon the
impedance in its plate circuit. Plate resistance is calculated from:
AEp
R
v
with Es and Esg constant.
4 -5
The
Mixer and Converter Tubes
superheterodyne receiver always in-
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80
Vacuum Tube
Principles
THE
RADIO
OSCILLATOR GRID
- PLATE
r_FSCREEN
CAT NODE
GRID
METAL SPELL
FILAMENT
`
SUPPRESSOR
AND SHELL
Figure
SIGNAL GRID
Figure 20
LEAD INDUCTANCE
GRID STRUCTURE OF 6SA7
The degenerative action of cathode lead inductance tends to reduce the effective grid-tocathode voltage with respect to the voltage
available across the input tuned circuit. Cathode lead inductance also introduces undesirable coupling between the input and the out-
CONVERTER TUBE
eludes at least one stage for changing the
frequency of the incoming signal to the fixed
frequency of the main intermediate amplifier
in the receiver. This frequency changing
process is accomplished by selecting the
beat -note difference frequency between a
locally generated oscillation and the incoming
signal frequency. If the oscillator signal is
supplied by a separate tube, the frequency
changing tube is called a mixer. Alternatively,
the oscillation may be generated by additional
elements within the frequency changer tube.
In this case the frequency changer is commonly called a converter tube.
Conversion
Conductance
The conversion conductance(Ge)
is a coefficient of interest in the
case of mixer or converter tubes,
or of conventional triodes, tetrodes, or pentodes operating as frequency changers. The
conversion conductance is the ratio of a
change in the signal -grid voltage at the input
frequency to a change in the output current at
the converted frequency. Hence Gc in a mixer
is essentially the same as transconductance
in an amplifier, with the exception that the
input signal and the output current are on different frequencies. The value of G, in conventional mixer tubes is from 300 to 1000
micromhos. The value of G, in an amplifier
tube operated as a mixer is approximately 0.3
the Gm of the tube operated as an amplifier.
The voltage gain of a mixer stage is equal to
GCZL where ZL is the impedance of the plate
load into which the mixer tube operates.
The simplest mixer tube is
the diode. The noise figure,
or figure of merit, for a mixer of this type is
not as good as that obtained with other more
complex mixers; however, the diode is useful
as a mixer in u -h -f and v -h -f equipment where
low interelectrode capacities are vital to circuit operation. Since the diode impedance is
The Diode Mixer
21
SHOWING THE EFFECT OF CATHODE
put circuits.
low, the local oscillator must furnish considerable power to the diode mixer. A good
diode mixer has an overall gain of about 0.5.
A triode mixer has better
gain and a better noise figure
than the diode mixer. At low frequencies, the
gain and noise figure of a triode mixer closely
approaches those figures obtained when the
tube is used as an amplifier. In the u -h -f and
v -h -f range, the efficiency of the triode mixer
deteriorates rapidly. The optimum local oscillator voltage for a triode mixer is about 0.7 as
large as the cutoff bias of the triode. Very
little local oscillator power is required by a
triode mixer.
The Triode Mixer
Pentode Mixers and
Converter Tubes
The most common multi grid converter tube for
broadcast or shortwave
use is the penta grid converter, typified by
the 6SA7, 6SB7 -Y and 6BA7 tubes (figure 20).
Operation of these converter tubes and pentode
mixers will be covered in the Receiver Fundamentals Chapter.
4 -6
Electron Tubes at Very
High Frequencies
As the frequency of operation of the usual
type of electron tube is increased above about
20 Mc., certain assumptions which are valid
for operation at lower frequencies must be reexamined. First, we find that lead inductances
from the socket connections to the actual
elements within the envelope no longer are
negligible. Second, we find that electron
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HANDBOOK
The
transit time no longer may be ignored; an
appreciable fraction of a cycle of input signal
may be required for an electron to leave the
cathode space charge, pass through the grid
wires, and travel through the space between
grid and plate.
The effect of lead inductance is two -fold. First, as
shown in figure 21, the
combination of grid -lead inductance, gridcathode capacitance, and cathode lead inductance tends to reduce the effective grid- cathode
signal voltage for a constant voltage at the
tube terminals as the frequency is increased.
Second, cathode lead inductance tends to
introduce undesired coupling between the
various elements within the tube.
Tubes especially designed for v -h -f and
u -h -f use have had their lead inductances
minimized. The usual procedures for reducing
lead inductance are: (1) using heavy lead
conductors or several leads in parallel (examples are the 6SH7 and 6AK5), (2) scaling
down the tube in all dimensions to reduce
both lead inductances and interelectrode
capacitances (examples are the 6AK5, 6F4,
and other acorn and miniature tubes), and (3)
the use of very low inductance extensions of
the elements themselves as external connections (examples are lighthouse tubes such as
the 2C40, oilcan tubes such as the 2C29, and
many types of v -h -f transmitting tubes).
Effects of
Lead Inductance
Effect of
Transit Time
When
erated
an electron tube is opat a frequency high
enough that electron transit
time between cathode and plate is an appreciable fraction of a cycle at the input frequency, several undesirable effects take place.
First, the grid takes power from the input
signal even though the grid is negative at all
times. This comes about since the grid will
have changed its potential during the time
required for an electron to pass from cathode
to plate. Due to interaction, and a resulting
phase difference between the field associated
with the grid and that associated with a moving electron, the grid presents a resistance to
an input signal in addition to its normal
"cold" capacitance. Further, as a result of
this action, plate current no longer is in phase
with grid voltage.
An amplifier stage operating at a frequency
high enough that transit time is appreciable:
(a) Is difficult to excite as a result of grid
loss from the equivalent input grid resistance,
(b) Is capable of less output since transconductance is reduced and plate current is
not in phase with grid voltage.
The effects of transit time increase with the
square of the operating frequency, and they
Klystron
81
increase rapidly as frequency is increased
above the value where they become just appreciable. These effects may be reduced by
scaling down tube dimensions; a procedure
which also reduces lead inductance. Further,
transit-time effects may be reduced by the
obvious procedure of increasing electrode potentials so that electron velocity will be increased. However, due to the law of electron motion in an electric field, transit time is
increased only as the square root of the ratio
of operating potential increase; therefore this
expedient is of limited value due to other
limitations upon operating voltages of small
electron tubes.
4 -7
Special Microwave
Electron Tubes
Due primarily to the limitation imposed by
transit time, conventional negative -grid electron tubes are capable of affording worthwhile
amplification and power output only up to a
definite upper frequency. This upper frequency
limit varies from perhaps 100 Mc. for conventional tube types to about 4000 Mc. for
specialized types such as the lighthouse tube.
Above the limiting frequency, the conventional
negative -grid tube no longer is practicable and
recourse must be taken to totally different
types of electron tubes in which electron
transit time is not a limitation to operation.
Three of the most important of such microwave
tube types are the klystron, the magnetron, and
the travelling wave tube.
The klystron is a type
of electron tube in which
electron transit time is used to advantage,
Such tubes comprise, as shown in figure 22,
a cathode, a focussing electrode, a resonator
connected to a pair of grids which afford
velocity modulation of the electron beam
(called the "buncher "), a drift space, and
another resonator connected to a pair of grids
(called the "catcher "). A collector for the
expended electrons may be included at the
end of the tube, or the catcher may also perform the function of electron collection.
The tube operates in the following manner:
The cathode emits a stream of electrons which
is focussed into a beam by the focussing
electrode. The stream passes through the
buncher where it is acted upon by any field
existing between the two grids of the buncher
cavity. When the potential between the two
grids is zero, the stream passes through without change in velocity. But when the potential
between the two grids of the buncher is increasingly positive in the direction of electron
The Power Klystron
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82
Vacuum
TWO- CAVITY
A conventional
with a feedback
two cavities so
Tube
Principles
Figure 22
KLYSTRON OSCILLATOR
two -cavity klystron is shown
loop connected between the
that the tube may be used as
an
oscillator.
motion, the velocity of the electrons in the
beam is increased. Conversely, when the field
becomes increasingly negative in the direction
of the beam (corresponding to the other half
cycle of the exciting voltage from that which
produced electron acceleration) the velocity
of the electrons in the beam is decreased.
When the velocity- modulated electron beam
reaches the drift space, where there is no field,
those electrons which have been sped up on
one half -cycle overtake those immediately
ahead which were slowed down on the other
half- cycle. In this way, the beam electrons become bunched together. As the bunched groups
pass through the two grids of the catcher
cavity, they impart pulses of energy to these
grids. The catcher grid -space is charged to
different voltage levels by the passing electron
bunches, and a corresponding oscillating field
is set up in the catcher cavity. The catcher is
designed to resonate at the frequency of the
velocity- modulated beam, or at a harmonic of
this frequency.
In the klystron amplifier, energy delivered
by the buncher to the catcher grids is greater
than that applied to the buncher cavity by the
input signal. In the klystron oscillator a feedback loop connects the two cavities. Coupling
to either buncher or catcher is provided by
small Loops which enter the cavities by way of
concentric lines.
The klystron is an electron -coupled device.
When used as an oscillator, its output voltage
is rich in harmonics. Klystron oscillators of
various types afford power outputs ranging
from less than I watt to many thousand watts.
Operating efficiency varies between 5 and 30
per cent. Frequency may be shifted to some
extent by varying the beam voltage. Tuning is
THE
RADIO
Figure 23
REFLEX KLYSTRON OSCILLATOR
A conventional reflex klystron oscillator of
the type commonly used as o local oscillator
in superheterodyne receivers operating above
about 2000 Mc. is shown above. Frequency
modulation of the output frequency of the oscillator, or o-f-c operation in a receiver, may be
obtained by varying the negative voltage on the
repeller electrode.
carried on mechanically in some klystrons by
altering (by means of knob settings) the shape
of the resonant cavity.
two -cavity klystron
as described in the preceding paragraphs is primarily used as a transmitting device since quite reasonable amounts
of power are made available in its output circuit. However, for applications where a much
smaller amount of power is required-power
levels in the milliwatt range for low -power
transmitters, receiver local oscillators, etc.,
another type of klystron having only a single
cavity is more frequently used.
The theory of operation of the single- cavity
klystron is essentially the same as the multi cavity type with the exception that the velocity- modulated electron beam, after having left
the " buncher" cavity is reflected back into
the area of the buncher again by a repeller
electrode as illustrated in figure 23. The
potentials on the various electrodes are adjusted to the value such that proper bunching
of the electron beam will take place just as a
particular portion of the velocity -modulated
beam reenters the area of the resonant cavity.
Since this type of klystron has only one circuit
it can be used only as an oscillator and not as
an amplifier. Effective modulation of the frequency of a single- cavity klystron for FM
work can be obtained by modulating the repeller electrode voltage.
The Reflex Klystron
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The
-
HANDBOOK
Magnetron
The
PLATE
83
MAGNET COIL
1
ANODE
ANODE
FIL
GRID
TERMINAL
ANODE
TERMINAL
II`
CATHODE
/
IL
ANODE
GLASS
`PLATE 2
SEAL
GLASS ENVELOPE
O
ANODE
OR'D
HEATER
FILAMENT
VOLTAGE
EYELET
SEAL
LEAD
TERMINAL
EYELET
TURULATiON
lower
Figure 24
CUTAWAY VIEW OF
WESTERN ELECTRIC 416- B/6280
VHF PLANAR TRIODE TUBE
The 416 -B, designed by the Bell
Telephone Laboratories is intended
for amplifier or frequency multiplier
service in the 4000 me region. Employing grid wires having a diameter
equal to fifteen wavelengths of light,
416 -B
has a transconductance of
50,000.
Spacing between grid and
cathode is .0005', to reduce transit
time effects. Entire tube is gold plated.
The Magnetron
magnetron is an
oscillator tube normally
The
PLATE
VOLTAGE
Figure 25
SIMPLE MAGNETRON OSCILLATOR
An external tank circuit is used with this type
of magnetron oscillator for operation in the
GLASS
the
FILAMENT
s -h -f
em-
ployed where very high values of peak power
or moderate amounts of average power are
required in the range from perhaps 700 Mc.
to 30,000 Mc. Special magnetrons were developed for wartime use in radar equipments
which had peak power capabilities of several
million watts (megawatts) output at frequencies in the vicinity of 3000 Mc. The normal
duty cycle of operation of these radar equipments was approximately 1 /10 of one per
cent (the tube operated about 1 /1000 of the
time and rested for the balance of the operating period) so that the average power output
of these magnetrons was in the vicinity of
1000 watts.
u -h
-f ronge.
In its simplest form the magnetron tube is a
filament -type diode with two half-cylindrical
plates or anodes situated coaxially with respect to the filament. The construction is
illustrated in figure 25A. The anodes of the
magnetron are connected to a resonant circuit
as illustrated on figure 25B. The tube is surrounded by an electromagnet coil which, in
turn, is connected to a low -voltage d -c energizing source through a rheostat R for controlling the strength of the magnetic field. The
field coil is oriented so that the lines of
magnetic force it sets up are parallel to the
axis of the electrodes.
Under the influence of the strong magnetic
field, electrons leaving the filament are deflected from their normal paths and move in
circular orbits within the anode cylinder. This
effect results in a negative resistance which
sustains oscillations. The oscillation frequency is very nearly the value determined by
L and C. In other magnetron circuits, the frequency may be governed by the electron rotation, no external tuned circuits being employed. Wavelengths of less than 1 centimeter have been produced with such circuits.
More complex magnetron tubes employ no
external tuned circuit, but utilize instead one
or more resonant cavities which are integral
with the anode structure. Figure 26 shows a
magnetron of this type having a multi -cellular
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84
Vacuum
Tube
Principles
-
RADIO
THE
WAVE GUIDE
CATNODE LEAOE
WAVE GU IDE
OUTPUT
INPUT
ELECTRON BEAM
MAGNETRON
PE MANE NT
CTNODE
MAGNET
ANODE ESSASS
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Figure 26
MODERN MULTI- CAVITY MAGNETRON
Illustrated is an external -anode strapped magnetron of the type commonly used in radar equipment for the 10 -cm. range. A permanent magnet
of the general type used with such a magnetron
Is shown in the right -hand portion of the drawing,
with the magnetron in place between the pole
pieces of the magnet.
anode of eight cavities. It will be noted, also,
that alternate cavities (which would operate at
the same polarity when the tube is oscillating)
are strapped together. Strapping was found to
improve the efficiency and stability of high power radar magnetrons. In most radar applications of magnetron oscillators a powerful
permanent magnet of controlled characteristics
is employed to supply the magnetic field
rather than the use of an electromagnet.
The Travelling
Wave Tube
Travelling Wave Tube
(figure 27) consists of a helix
located within an evacuated
envelope. Input and output terminations are
affixed to each end of the helix. An electron
beam passes through the helix and interacts
with a wave travelling along the helix to produce broad band amplification at microwave
frequencies.
When the input signal is applied to the gun
end of the helix, it travels along the helix wire
at approximately the speed of light. However,
the signal velocity measured along the axis
of the helix is considerably lower. The electrons emitted by the cathode gun pass axially
through the helix to the collector, located at
the output end of the helix. The average velocity of the electrons depends upon the potential
of the collector with respect to the cathode.
When the average velocity of the electrons is
greater than the velocity of the helix wave,
the electrons become crowded together in the
various regions of retarded field, where they
impart energy to the helix wave. A power gain
of 100 or more may be produced by this tube.
4 -8
The
The Cathode -Ray Tube
The Cathode -Ray Tube
cathode -ray
The
is
a
tube
special type of
ANODE
COLLECTOR
Figure 27
THE TRAVELLING WAVE TUBE
Operation of this tube is the result of inter.
action between the electron beam and wave
travelling along the helix.
electron tube which permits the visual observation of electrical signals. It may be incorporated into an oscilloscope for use as a test
instrument or it may be the display device for
radar equipment or a television receiver.
cathode -ray tube always includes an electron gun for producing a stream of electrons, a
grid for controlling the intensity of the electron beam, and a luminescent screen for converting the impinging electron beam into visible light. Such a tube always operates in conjunction with either a built -in or an external
means for focussing the electron stream into a
narrow beam, and a means for deflecting the
electron beam in accordance with an electrical
signal.
The main electrical difference between
types of cathode -ray tubes lies in the means
employed for focussing and deflecting the
electron beam. The beam may be focussed
and/or deflected either electrostatically or
magnetically, since a stream of electrons can
be acted upon either by an electrostatic or a
magnetic field. In an electrostatic field the
electron beam tends to be deflected toward the
positive termination of the field (figure 28).
In a magnetic field the stream tends to be
deflected at right angles to the field. Further,
an electron beam tends to be deflected so that
it is normal (perpendicular) to the equipotential
lines of an electrostatic field- and it tends to
be deflected so that it is parallel to the lines
of force in a magnetic field.
Large cathode -ray tubes used as kinescopes
in television receivers usually are both focused
and deflected magnetically. On the other hand,
the medium -size CR tubes used in oscilloscopes and small television receivers usually
are both focused and deflected electrostatically. But CR tubes for special applications
may be focused magnetically and deflected
electrostatically or vice versa.
There are advantages and disadvantages to
Operation of
the CRT
A
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HANDBOOK
The
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Figure 28
TYPICAL ELECTROSTATIC
CATHODE -RAY TUBE
both types of focussing and deflection. However, it may be stated that electrostatic deflection is much better than magnetic deflection
when high -frequency waves are to be displayed
on the screen; hence the almost universal use
of this type of deflection for oscillographic
work. But when a tube is operated at a high
value of accelerating potential so as to obtain
a bright display on the face of the tube as for
television or radar work, the use of magnetic
deflection becomes desirable since it is relatively easier to deflect a high -velocity electron
than electrostatically.
beam magnetically
However, an ion trap is required with magnetic deflection since the heavy negative ions
emitted by the cathode are not materially deflected by the magnetic field and hence would
burn an "ion spot" in the center of the luminescent screen. With electrostatic deflection
the heavy ions are deflected equally as well
as the electrons in the beam so that an ion
spot is not formed.
Construction of
The construction of a typical
Electrostatic CRT electrostatic- focus, electrostatic- deflection cathode-ray
tube is illustrated in the pictorial diagram of
figure 28. The indirectly heated cathode K releases free electrons when heated by the
enclosed filament. The cathode is surrounded
by a cylinder G, which has a small hole in its
front for the passage of the electron stream.
Although this element is not a wire mesh as
is the usual grid, it is known by the same
name because its action is similar: it controls
the electron stream when its negative potential
is varied.
Next in order, is found the first accelerating
anode, H, which resembles another disk or
cylinder with a small hole in its center. This
electrode is run at a high or moderately high
positive voltage, to accelerate the electrons
towards the far end of the tube.
The focussing electrode, F, is a sleeve
which usually contains two small disks, each
with a small hole.
After leaving the focussing electrode, the
electrons pass through another accelerating
Cathode
Ray
Tube
85
anode, A, which is operated at a high positive
potential. In some tubes this electrode is operatcd at a higher potential than the first accelerating electrode, H, while in other tubes both
accelerating electrodes are operated at the
same potential.
The electrodes which have been described
this point constitute the electron gun,
which produces the free electrons and focusses
them into a slender, concentrated, rapidly traveling stream for projecting onto the viewing screen.
up to
Electrostatic
Deflection
To make the tube useful, means
must be provided for deflecting
the electron beam along two axes
at right angles to each other. The more corn mon tubes employ electrostatic deflection
plates, one pair to exert a force on the beam
in the vertical plane and one pair to exert a
force in the horizontal plane. These plates
are designated as B and C in figure 28.
Standard oscilloscope practice with small
cathode -ray tubes calls for connecting one of
the B plates and one of the C plates together
and to the high voltage accelerating anode.
With the newer three -inch tubes and with five inch tubes and larger, all four deflecting plates
are commonly used for deflection. The positive
high voltage is grounded, instead of the negative as is common practice in amplifiers, etc.,
in order to permit operation of the deflecting
plates at a d-c potential at or near ground.
An Aquadag coating is applied to the inside
of the envelope to attract any secondary electrons emitted by the flourescent screen.
In the average electrostatic -deflection CR
tube the spot will be fairly well centered if all
four deflection plates are returned to the po-
tential of the second anode (ground). How ever, for accurate centering and to permit moving the entire trace either horizontally or
vertically to permit display of a particular
waveform, horizontal and vertical centering
controls usually are provided on the front of
the oscilloscope.
After the spot is once centered, it is necessary only to apply a positive or negative voltage (with respect to ground) to one of the
ungrounded or "free" deflector plates in order
to move the spot. If the voltage is positive
with respect to ground, the beam will be
attracted toward that deflector plate, while if
negative the beam and spot will be repulsed.
The amount of deflection is directly proportional to the voltage (with respect to ground)
that is applied to the free electrode.
With the larger- screen higher -voltage tubes
it becomes necessary to place deflecting voltage on both horizontal and both vertical plates.
This is done for two reasons: First, the amount
of deflection voltage required by the high-
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Vacuum
86
Tube
FIRST
ANODE
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THE
DEFLECTION COILS
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Figure 29
TYPICAL ELECTROMAGNETIC
CATHODE -RAY TUBE
voltage tubes is so great that a transmitting
tube operating from a high voltage supply
would be required to attain this voltage without distortion. By using push -pull deflection
with two tubes feeding the deflection plates,
the necessary plate supply voltage for the deflection amplifier is halved. Second, a certain
amount of de- focussing of the electron stream
is always present on the extreme excursions in
deflection voltage when this voltage is applied
only to one deflecting plate. When the deflecting voltage is fed in push -pull to both
deflecting plates in each plane, there is no defocussing because the average voltage acting
on the electron stream is zero, even though the
net voltage (which causes the deflection)
acting on the stream is twice that on either
plate.
The fact that the beam is deflected by a
magnetic field is important even in an oscilloscope which employs a tube using electrostatic deflection, because it means that precautions must be taken to protect the tube from
the transformer fields and sometimes even the
earth's magnetic field. This normally is done
by incorporating a magnetic shield around the
tube and by placing any transformers as far
from the tube as possible, oriented to the position which produces minimum effect upon the
electron stream.
The
electromagnetic
cathode -ray tube allows
greater definition than
does the electrostatic tube. Also, electromagnetic definition has a number of advantages when a rotating radial sweep is required
to give polar indications.
The production of the electron beam in an
electromagnetic tube is essentially the same
as in the electrostatic tube. The grid structure
is similar, and controls the electron beam in
an identical manner. The elements of a typical
electromagnetic tube are shown in figure 29.
The focus coil is wound on an iron core which
may be moved along the neck of the tube to
focus the electron beam. For final adjustment,
Construction of Electro-
magnetic CRT
.1m
Jew
.1n1R
Figure 30
Two pairs of coils arranged for electromagnetic deflection in two directions.
the current flowing in the coil may be varied.
A second pair of coils, the deflection coils
are mounted at right angles to each other
around the neck of the tube. In some cases,
these coils can rotate around the axis of the
tube.
Two anodes are used for accelerating the
electrons from the cathode to the screen. The
second anode is a graphite coating (Aquadag)
on the inside of the glass envelope. The function of this coating is to attract any secondary
electrons emitted by the flourescent screen,
and also to shield the electron beam.
In some types of electromagnetic tubes, a
first, or accelerating anode is also used in
addition to the Aquadag.
Electromagnetic
Deflection
magnetic field will deflect
an electron beam in a direction which is at right angles
to both the direction of the field and the direction of motion of the beam.
In the general case, two pairs of deflection
coils are used (figure 30). One pair is for
horizontal deflection, and the other pair is for
vertical deflection. The two coils in a pair
are connected in series and are wound in such
directions that the magnetic field flows from
one coil, through the electron beam to the
other coil. The force exerted on the beam by
the field moves it to any point on the screen
by application of the proper currents to these
A
coils.
The human eye retains an image
for about one - sixteenth second
after viewing. In a CRT, the spot can be
moved so quickly that a series of adjacent
spots can be made to appear as a line, if the
beam is swept over the path fast enough. As
The Trace
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HANDBOOK
Gas
long as the electron beam strikes in a given
place at least sixteen times a second, the
spot will appear to the human eye as a source
of continuous light with very little flicker.
-
least five types of luminescent screen materials
are commonly available on
the various types of CR tubes commercially
available. These screen materials are called
phosphors; each of the five phosphors is best
suited to a particular type of application. The
P -1 phosphor, which has a green flourescence
with medium persistence, is almost invariably
used for oscilloscope tubes for visual observaScreen Materials
"Phosphors"
At
tion. The P -4 phosphor, with white fluorescence and medium persistence, is used on
television viewing tubes ( "Kinescopes "). The
P -5 and P -11 phosphors, with blue fluorescence and very short persistence, are used
primarily in oscilloscopes where photographic
recording of the trace is to be obtained. The
P -7 phosphor, which has a blue flash and a
long -persistence greenish -yellow persistence,
is used primarily for radar displays where
retention of the image for several seconds
after the initial signal display is required.
4 -9
Gas Tubes
The space charge of electrons in the vicinity
of the cathode in a diode causes the plate -tocathode voltage drop to be a function of the
current being carried between the cathode and
the plate. This voltage drop can be rather high
when large currents are being passed, causing
a considerable amount of energy loss which
shows up as plate dissipation.
Mercury Vapor
Tubes
Tubes
87
Mercury -vapor tubes, although
very widely used, have the
disadvantage that they must be
operated within a specific temperature range
(25° to 70°C.) in order that the mercury vapor
pressure within the tube shall be within the
proper range. If the temperature is too low,
the drop across the tube becomes too high
causing immediate overheating and possible
damage to the elements. If the temperature is
too high, the vapor pressure is too high, and
the voltage at which the tube will "flash back"
is lowered to the point where destruction of
the tube may take place. Since the ambient
temperature range specified above is within
the normal room temperature range, no trouble
will be encountered under normal operating
conditions. However, by the substitution of
xenon gas for mercury it is possible to produce a rectifier with characteristics comparable
to those of the mercury -vapor tube except that
the tube is capable of operating over the range
from approximately -70° to 90° C. The 3B25
rectifier is an example of this type of tube.
If a grid is inserted between the cathode and plate of a mercury -vapor
gaseous- conduction rectifier, a negative potential placed upon the added element
will increase the plate -to- cathode voltage drop
required before the tube will ionize or "fire."
The potential upon the control grid will have
no effect on the plate -to- cathode drop after the
tube has ionized. However, the grid voltage
may be adjusted to such a value that conduction will take place only over the desired
portion of the cycle of the a-c voltage being
impressed upon the plate of the rectifier.
Thyratron
Tubes
Voltage Regulator In a glow -discharge gas tube
Tubes
the voltage drop across the
electrodes remains constant
The negative space charge can
be neutralized by the presence
of the proper density of positive
ions in the space between the cathode and
anode. The positive ions may be obtained by
the introduction of the proper amount of gas or
a small amount of mercury into the envelope of
the tube. Then the voltage drop across the
tube reaches the ionization potential of the
gas or mercury vapor, the gas molecules will
become ionized to form positive ions. The
positive ions then tend to neutralize the space
charge in the vicinity of the cathode. The voltage drop across the tube then remains constant
at the ionization potential of the gas up to a
current drain equal to the maximum emission
capability of the cathode. The voltage drop
varies between 10 and 20 volts, depending
upon the particular gas employed, up to the
maximum current rating of the tube.
Action of
Positive Ions
over a wide range of current passing through
the tube. This property exists because the
degree of ionization of the gas in the tube
varies with the amount of current passing
through the tube. When a large current is
passed, the gas is highly ionized and the
internal impedance of the tube is low. When a
small current is passed, the gas is lightly
ionized and the internal impedance of the tube
is high. Over the operating range of the tube,
the product (IR) of the current through the tube
and the internal impedance of the tube is very
nearly constant. Examples of this type of tube
are VR -150, VR -105 and the old 874.
Vacuum tubes are grouped into
three major classifications:
commercial, ruggedized, and
premium (or reliable). Any one of these three
groups may also be further classified for
Vacuum Tube
Classification
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88
Tube
Vacuum
THE
Principles
100-
military duty (JAN classification). To qualify
for JAN classification, sample lots of the
particular tube must have passed special
qualification tests at the factory. It should not
be construed that a JAN-type tube is better
than a commercial tube, since some commercial
tests and specifications are more rigid than
the corresponding JAN specifications. The
JAN -stamped tube has merely been accepted
under a certain set of conditions for military
service.
IP=2.5IAA.
eo
60
L
0
20
0
0
10
20
EP
Ruggedized or
Premium Tubes
Radio tubes are being used in
increasing numbers for industrial applications, such as
computing and control machinery, and in aviation and marine equipment. When a tube fails
in a home radio receiver, it is merely inconvenient, but a tube failure in industrial applications may bring about stoppage of some vital
process, resulting in financial loss, or even
danger to life.
To meet the demands of these industrial
applications, a series of tubes was evolved
incorporating many special features designed
to ensure a long and pre- determined operating
life, and uniform characteristics among similar
tubes. Such tubes are known as ruggedized or
premium tubes. Early attempts to select reTRIODE PLATE
,
`FLUORESCENT ANODE
TRIODE GRID
RAY CONTROL
ELECTRODE
CATHODES
Figure 31
SCHEMATIC REPRESENTATION
OF "MAGIC EYE" TUBE
liable specimens of tubes from ordinary stock
tubes proved that in the long run the selected
tubes were no better than tubes picked at
random. Long life and ruggedness had to be
built into the tubes by means of proper choice
and 100% inspection of all materials used in
the tube, by critical processing inspection and
assembling, and by conservative ratings of the
tube.
Pure tungsten wire is used for heaters in
preference to alloys of lower tensile strength.
Nickel tubing is employed around the heater
wires at the junction to the stem wires to
reduce breakage at this point. Element structures are given extra supports and bracing.
Finally, all tubes are given a 50 hour test run
under full operating conditions to eliminate
early failures. When operated within their
ratings, ruggedized or premium tubes should
provide a life well in excess of 10,000 hours.
Ruggedized tubes will withstand severe
impact shocks for short periods, and will
RADIO
30
40
VOLTS)
50
60
Figure 32
AMPLIFICATION FACTOR OF TYPICAL MODE
TUBE DROPS RAPIDLY AS PLATE VOLTAGE
IS
DECREASED BELOW 20 VOLTS
operate under conditions of vibration for many
hours. The tubes may be identified in many
cases by the fact that their nomenclature includes a "W" in the type number, as in 807W,
5U4W, etc. Some ruggedized tubes are included
in the "5000" series nomenclature. The 5654
is a ruggedized version of the 6AK5, the 5692
is a ruggedized version of the 6SN7, etc.
4 -10
lectron
Miscellaneous Tube Types
electron -ray tube or magic eye
contains two sets of elements, one
of which is a triode amplifier and
the other a cathode -ray indicator. The plate of
the triode section is internally connected to
the ray- control electrode (figure 31), so that
as the plate voltage varies in accordance with
the applied signal the voltage on the ray -control
electrode also varies. The ray -control electrode
is a metal cylinder so placed relative to the
cathode that it deflects some of the electrons
emitted from the cathode. The electrons which
strike the anode cause it to fluoresce, or give
off light, so that the deflection caused by the
ray -control electrode, which prevents electrons
from striking part of the anode, produces a
wedge- shaped electrical shadow on the fluorescent anode. The size of this shadow is determined by the voltage on the ray -electrode. When
this electrode is at the same potential as the
fluorescent anode, the shadow disappears; if
the ray -electrode is less positive than the
anode, a shadow appears the width of which
is proportional to the voltage on the ray -electrode. Magic eye tubes may be used as tuning
indicators, and as balance indicators in electrical bridge circuits. If the angle of shadow is
calibrated, the eye tube may be used as a voltmeter where rough measurements suffice.
E
The
Ray Tubes
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Miscellaneous
HANDBOOK
89
Ecz (VOLTS)
Ec1=,z.ev
Figure 33
CHARACTERISTIC CURVES OF 12AK5
SFACE- CHARGE TRIODE
Controlled
Warm -up
Tubes
Series heater strings are employed
in ac -dc radio receivers and television sets to reduce the cost,
size, and weight of the equipment.
Voltage surges of great magnitude occur in
series operated filaments because of variations
in the rate of warm -up of the various tubes.
As the tubes warm up, the heater resistance
changes. This change is not the same between
tubes of various types, or even between tubes
same type made by different manufacturers. Some 6 -volt tubes show an initial
surge as high as 9 -volts during warm -up, while
slow -heating tubes such as the 25BQ6 are
underheated during the voltage surge on the
6 -volt tubes.
the
of
Standardization of heater characteristics in
a new group of cubes designed for series heater
strings has eliminated this trouble. The new
tubes have either 600 ma. or 400 ma. heaters,
with a controlled warm -up time of approximately
11 seconds. The 5U8, 6CG7, and 12BH7 -A are
examples of controlled warm -up tubes.
Introduction of the 12 -volt ignition
system in American automobiles
Potential
has brought about the design of a
Tubes
series of tubes capable of operation
with a plate potential of 12 -14
volts. Standard tubes perform poorly at low
plate potentials, as the amplification factor
of the tube drops rapidly as the plate voltage
is decreased (figure 32). Contact potential
effects, and change of characteristics with
variations of filament voltage combine to make
operation at low plate potentials even more
Low
Plate
erratic.
By employing
and by altering the electrode geometry a series
of low voltage tubes has been developed by
Tung -Sol that effectively perform with all electrodes energized by a 12 -volt system. With a
suitable power output transistor, this makes
possible an automobile radio without a vibrator
power supply. A special space- charge tube
(12K5) has been developed that delivers 40
milliwatts of audio power with a 12 volt plate
supply (figure 33).
The increased number of imported
radios and high- fidelity equipment
have brought many foreign vacuum
tubes into the United States. Many of these
tubes are comparable to, or interchangeable
with standard American tubes. A complete
listing of the electrical characteristics and
hase connection diagrams of all general purpose tubes made in all tube -producing
countries outside the "Iron Curtain" is contained in the Radio Tube Vade Mecum (World's
Radio Tubes) available at most larger radio
parts jobbers or by mail from the publishers
of this Handbook. The Equivalent Tubes Vade
Mecum (World's Equivalent Tubes) gives all
replacement tubes for a given type, both exact
and near -equivalents (with points of difference detailed). (Data on TV and special purpose tubes if needed is contained in a companion volume Television Tubes Vade Mecum).
Foreign
Tubes
special processing techniques
www.americanradiohistory.com
CHAPTER FIVE
Transistors and
Semi -Conductors
One of the earliest detection devices used
in radio was the galena crystal, a crude example of a semiconductor. More modern examples of semiconductors are the copper -
5 -1
It has been previously stated that the electrons in an element having a large atomic
number are grouped into rings, each ring having a definite number of electrons. Atoms in
which these rings are completely filled are
called inert gases, of which helium and argon
are examples. All other elements have one or
more incomplete rings of electrons. If the incomplete ring is loosely bound, the electrons
may be easily removed, the element is called
metallic, and is a conductor of electric current.
If the incomplete ring is tightly bound, with
only a few missing electrons, the element is
called non - metallic and is an insulator of electric current. Germanium and silicon fall between these two sharply defined groups, and
exhibit both metallic and non -metallic characteristics. Pure germanium or silicon may be
considered to be a good insulator. The addition
of certain impurities in carefully controlled
amounts to the pure germanium will alter the
conductivity of the material. In addition, the
choice of the impurity can change the direction
of conductivity through the crystal, some impurities increasing conductivity to positive voltages, and others increasing conductivity to negative voltages.
oxide rectifier, the selenium rectifier and the
germanium diode. All of these devices offer
the interesting property of greater resistance
to the flow of electrical current in one direction than in the opposite direction. Typical
conduction curves for these semiconductors
are shown in Figure 1. The copper oxide rectifier action results from the function of a thin
film of cuprous oxide formed upon a pure copper disc. This film offers low resistance for
positive voltages, and high resistance for
negative voltages. The same action is observed in selenium rectifiers, where a film of
selenium is deposited on an iron surface.
s
1
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CD0IIT*L DIODI
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TYPICAL STATIC CHARACTERISTICS
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Mechanism of
Conduction
As indicated by their name, semiconductors
are substances which have a conductivity
intermediate between the high values observed
for metals and the low values observed for insulating materials. The mechanism of conduction in semiconductors is different from that
a
VOLTS
Figure
Atomic Structure of
Germanium and Silicon
lA
TYPICAL CHARACTERISTIC CURVE
OF SEMI -CONDUCTOR DIODE
90
www.americanradiohistory.com
Transistors
observed in metallic conductors. There exist
in semiconductors both negatively charged
electrons and positively charged particles,
called holes, which behave as though they
had a positive electrical charge equal in magnitude to the negative electrical charge on
the electron. These holes and electrons drift
in an electrical field with a velocity which is
proportional to the field itself:
SCHEMATIC REPRESENTATION
VAN
where
VAN
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ANODES
Color
CATB000
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Calorr Bao ft
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TUBE. GERMANIUM. SILICON
AND SELENIUM DIODES
Figure
COMMON DIODE
AND MARKINGS
IN ABOVE
1
91
-B
COLOR CODES
ARE SHOWN
CHART
=
µnE
= drift velocity of hole
= magnitude of electric field
= mobility of hole
In an electric field the holes will drift in a
direction opposite to that of the electron and
with about one-half the velocity, since the
hole mobility is about one -half the electron
mobility. A sample of a semiconductor, such as
germanium or silicon, which is both chemically
pure and mechanically perfect will contain in it
approximately equal numbers of holes and electrons and is called an intrinsic semiconductor.
The intrinsic resistivity of the semiconductor
depends strongly upon the temperature, being
about 50 ohm /cm. for germanium at room
temperature. The intrinsic resistivity of silicon
is about 65,000 ohm /cm. at the same temperature.
If, in the growing of the semiconductor crystal, a small amount of an impurity, such as
phosphorous, arsenic or antimony is included
in the crystal, each atom of the impurity contributes one free electron. This electron is
available for conduction. The crystal is said
to be doped and has become electron- conductPe- Nb JUNCTION
PLASTIC CASE
b-PC JUNCTION
P - TYPE
N- TYPE
GERMANIUM
CRYSTAL LAYER
GERMANIUM
CRYSTAL LAYER
COLLECTOR
EMITTER
LI
Nb
P
.MS'
BASE CONNECTION
SMALL 3 -PIN
BASE
L
Jw
y1!
o
o
Pt
4-
Z
.320
ASE CONNECTION
EMITTER
COLLECTOR
Figure 2A
CUT -AWAY VIEW OF JUNCTION
TRANSISTOR, SHOWING PHYSICAL
ARRANGEMENT
Figure 2B
PICTORIAL EQUIVALENT OF
P -N -P JUNCTION TRANSISTOR
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SIGN Z
92
THE RADIO
Transistors and Semi- Conductors
EMITTER
COLLECTOR
EMITTER
COLLECTOR
BASE-I
BISE
TRANSISTOR OR
POINT CONTACT TRANSISTOR
N-P-N TRANSISTOR
P -N -P
Figure 4
ELECTRICAL SYMBOLS
FOR TRANSISTORS
Figure 3
CONSTRUCTION DETAIL OF A
POINT CONTACT TRANSISTOR
ing in nature and is called N (negative) type
germanium. The impurities which contribute
electrons are called donors. N -type germanium
has better conductivity than pure germanium in
one direction, and a continuous stream of electrons will flow through the crystal in this direction as long as an external potential of the
correct polarity is applied across the crystal.
Other impurities, such as aluminum, gallium or indium add one hole to the semiconducting crystal by accepting one electron for
each atom of impurity, thus creating additional
holes in the semiconducting crystal. The material is now said to be hole- conducting, or P
(positive) type germanium. The impurities
which create holes are called acceptors. P -type
germanium has better conductivity than pure
germanium in one direction. This direction is
opposite to that of the N -type material. Either
the N -type or the P -type germanium is called
extrinsic conducting type. The doped materials
have lower resistivities than the pure materials,
and doped semiconductors in the resistivity
range of .01 to 10 ohm /cm. are normally used
in the production of transistors.
5-3
The Transistor
In the past few years an entire new technology has been developed for the application
of certain semiconducting materials in production of devices having gain properties. These
gain properties were previously found only in
vacuum tubes. The elements germanium and
silicon are the principal materials which exhibit the proper semiconducting properties permitting their application in the new amplifying devices called transistors. However,
other semiconducting materials, including the
compounds indium antimonide and lead sulfide
have been used experimentally in the production of transistors.
Types of Transistors
There
are two basic
types of transistors, the
point-contact type and the junction type (figure 2) . Typical construction detail of a pointcontact transistor is shown in Figure 3, and
the electrical symbol is shown in Figure 4. The
emitter and collector electrodes make contact
with a small block of germanium, called the
base. The base may be either N -type or P -type
germanium, and is approximately .05" long
and .03" thick. The emitter and collector electrodes are fine wires, and are spaced about
.005" apart on the germanium base. The complete assembly is usually encapsulated in a
small, plastic case to provide ruggedness and
to avoid contaminating effects of the atmosphere. The polarity of emitter and collector
voltages depends upon the type of germanium
employed in the base, as illustrated in figure 4.
The junction transistor consists of a piece
of either N -type or P -type germanium between
two wafers of germanium of the opposite type.
Either N -P -N or P -N -P transistors may be
made. In one construction called the grown
crystal process, the original crystal, grown
from molten germanium or silicon, is created
in such a way as to have the two closely spaced
junctions imbedded in it. In the other construction called the fusion process, the crystals
are grown so as to make them a single conductivity type. The junctions are then produced by fusing small pellets of special metal
alloys into minute plates cut from the original
crystal. Typical construction detail of a junction
transistor is shown in figure 2A.
The electrical schematic for the P -N -P junction transistor is the same as for the pointcontact type, as is shown in figure 4.
Transistor Action
Presently available types of
transistors have three essential actions which collectively are called
transistor action. These are: minority carrier
injection, transport, and collection. Figure 2B
shows a simplified drawing of a P-N -P junction -type transistor, which can illustrate this
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HANDBOOK
Transistors
collective action. The P -N -P transistor consists of a piece of N -type germanium on opposite sides of which a layer of P -type material has been grown by the fusion process.
Terminals are connected to the two P- sections
and to the N -type base. The transistor may be
considered as two P -N junction rectifiers
p!aced in close juxaposition with a semi conduction crystal coupling the two rectifiers
together. The left -hand terminal is biased in
the forward (or conducting) direction and is
called the emitter. The right -hand terminal is
biased in the back (or reverse) direction and
is called the collector The operating potentials
are chosen with respect to the base terminal,
which may or may not be grounded. If an
N -P -N transistor is used in place of the P -N -P,
the operating potentials are reversed.
The P.
Nb junction on the left is biased
in the forward direction and holes from the
P region are injected into the Nb region, producing therein a concentration of holes substantially greater than normally present in the
material. These holes travel across the base
region towards the collector, attracting neighboring electrons, finally increasing the available supply of conducting electrons in the
collector loop. As a result, the collector loop
possesses lower resistance whenever the emitter circuit is in operation. In junction transistors this charge transport is by means of
diffusion wherein the charges move from a
region of high concentration to a region of
lower concentration at the collector. The collector, biased in the opposite direction, acts
as a sink for these holes, and is said to collect them.
It is known that any rectifier biased in the
forward direction has a very low internal impedance, whereas one biased in the back direction has a very high internal impedance. Thus,
current flows into the transistor in a low impedance circuit, and appears at the output as
current flowing in a high impedance circuit.
The ratio of a change in collector current to
a change in emitter current is called the current
amplification, or alpha:
-
a
=
ie
= current amplification
= change in collector current
i. = change in emitter current
where a
ie
Values of alpha up to 3 or so may be obtained in commercially available point- contact
transistors, and values of alpha up to about
0.95 are obtainable in junction transistors.
93
Alpha Cutoff
The alpha cutoff frequency of
a transistor is that frequency
at which the grounded base
current gain has decreased to 0.7 of the gain
obtained at 1 kc. For audio transistors, the
alpha cutoff frequency is in the region of 0.7
Mc. to 1.5 Mc. For r -f and switching transistors, the alpha cutoff frequency may be 5 Mc.
or higher. The upper frequency limit of operation of the transistor is determined by the
small but finite time it takes the majority carrier to move from one electrode to another.
Frequency
Drift Transistors
As previously noted, the
signal current in a conventional transistor is transmitted across the
base region by a diffusion process. The transit
time of the carriers across this region is, therefore relatively long. RCA has developed a
technique for the manufacture of transistors
which does not depend upon diffusion for
transmission of the signal across the base region. Transistors featuring this new process are
known as drift transistors. Diffusion of charge
carriers across the base region is eliminated and
the carriers are propelled across the region by
a "built in electric field. The resulting reduction of transit time of the carrier permits drift
transistors to be used at much higher frequencies than transistors of conventional design.
The "built in" electric field is in the base
region of the drift transistor. This field is
achieved by utilizing an impurity density
which varies from one side of the base to the
other. The impurity density is high next to
the emitter and low next to the collector. Thus,
there are more mobile electrons in the region
near the emitter than in the region near the
collector, and they will try to diffuse evenly
throughout the base. However, any displacement of the negative charge leaves a positive
charge in the region from which the electrons
came, because every atom of the base material
was originally electrically neutral. The displacement of the charge creates an electric
field that tends to prevent further electron diffusion so that a condition of equilibrium is
reached. The direction of this field is such as
to prevent electron diffusion from the high
density area near the emitter to the low density
area near the collector. Therefore, holes entering the base will be accelerated from the emitter to the collector by the electric field. Thus
the diffusion of charge carriers across the base
region is augmented by the built -in electric
field. A potential energy diagram for a drift
transistor is shown in figure 5.
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94
\
%
ri_
\
:i1,
r_=/
THE RADIO
Transistors and Semi -Conductors
DECREASING
POTENTIAL ENERGY
OF
S
MAJORITY CARRIER
h
WW
7
CC
6
34
REGION
u
I
EMITTER
CASE
REGION
I
I
ION
I
COLLECTOR
REGION
I
Ifi/Ciioill111E.1
IMTINEE:011101
o1 l'/.
'/_!=
S
DRIFT
u
M
IMI
.
,1\I/I
IIIPAlf
I/ICJ
1
10
I
I
DISTANCE
Figure
20
s0
40
50
COLLECTOR VOLTS
0
5
POTENTIAL ENERGY DIAGRAM
DRIFT TRANSISTOR (2N247)
FOR
5 -4
_Aiii/1111
Transistor
Characteristics
The transistor produces results that may be
comparable to a vacuum tube, but there is a
basic difference between the two devices. The
vacuum tube is a voltage controlled device
whereas the transistor is a current controlled
device. A vacuum tube normally operates with
its grid biased in the negative or high resistance direction, and its plate biased in the
positive or low resistance direction. The tube
conducts only by means of electrons, and has
its conducting counterpart in the form of the
N -P -N transistor, whose majority carriers are
also electrons. There is no vacuum tube equivalent of the P -N -P transistor, whose majority
carriers are holes.
The biasing conditions stated above provide
the high input impedance and low output impedance of the vacuum tube. The transistor is
biased in the positive or low resistance direction in the emitter circuit, and in the negative,
or high resistance direction in the collector
circuit resulting in a low input impedance
and a high output impedance, contrary to and
opposite from the vacuum tube. A comparison
of point-contact transistor characteristics and
vacuum tube characteristics is made in figure 6.
The resistance gain of a transistor is expressed as the ratio of output resistance to
input resistance. The input resistance of a
typical transistor is low, in the neighborhood
of 300 ohms, while the output resistance is
relatively high, usually over 20,000 ohms. For
a point- contact transistor, the resistance gain
is usually over 60.
The voltage gain of a transistor is the
product of alpha times the resistance gain,
and for a point- contact transistor is of the
FAlBM!lPA
25
so
75
100
125
ISO
175
200
PLATE VOLTS
Figure 6
COMPARISON OF POINT -CONTACT
TRANSISTOR AND VACUUM TUBE
CHARACTERISTICS
order of 3 X 60 = 180. A junction transistor
which has a value of alpha less than unity
nevertheless has a resistance gain of the order
of 2000 because of its extremely high output
resistance, and the resulting voltage gain is
about 1800 or so. For both types of transistors
the power gain is the product of alpha squared
times the resistance gain and is of the order
of 400 to 500.
The output characteristics of the junction
transistor are of great interest. A typical example is shown in figure 7. It is seen that the
junction transistor has the characteristics of
an ideal pentode vacuum tube. The collector
current is practically independent of the collector voltage. The range of linear operation
extends from a minimum voltage of about 0.2
volts up to the maximum rated collector voltage. A typical load line is shown, which illustrates the very high load impedance that
would be required for maximum power transfer. A grounded emitter circuit is usually used,
since the output impedance is not as high as
when a grounded base circuit is used.
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HANDBOOK
Transistor Characteristics
d
95
le
COLLECTOR
EMITTER
CASE
VALUES
OF THE EQUIVALENT
CIRCUIT
POINT- CONTACT
-1 0 -0.S
0
+5
+10
+ 5
+20
ISTOR
+25
COLLECTOR VOLTS
Vs..iMA VC15V.)
The output characteristics of a typical point contact transistor are shown in figure 6. The
pentode characteristics are less evident, rind the
output impedance is much lower, with the
range of linear operation extending down to
a collector voltage of 2 or 3. Of greater practical interest, however, is the input characteristic curve with short -circuited, or nearly shortcircuited input, as shown in figure 8. It is
this point -contact transistor characteristic of
having a region of negative impedance that
lends the unit to use in switching circuits. The
transistor circuit may be made to have two,
one or zero stable operating points, depending
upon the bias voltages and the load impedance
used.
Equivalent Circuit
of a Transistor
As is known from net -
work theory, the small
signal performance of
any device in any network can be represented
by means of an equivalent circuit. The most
EMITTER MILLIAMPERES
(te)
Figure 8
EMITTER CHARACTERISTIC CURVE
FOR TYPICAL POINT CONTACT
TRANSISTOR
JUNCTION
ISTOR
IMA VCR SV.)
re -EMITTER
1O0ß
SOA
Cb -SASE
300A
SOOA
RESISTANCE
Figure 7
OUTPUT CHARACTERISTICS OF
TYPICAL JUNCTION TRANSISTOR
(LE
RESISTANCE
RESCSÁL10EOR
c4- CURRENT
AMPLIFICATION
20000A
2.0
1
MEGONM
0.57
Figure 9
LOW FREQUENCY EQUIVALENT
(Common Bose) CIRCUIT FOR POINT
CONTACT AND JUNCTION
TRANSISTOR
convenient equivalent circuit for the low frequency small signal performance of both pointcontact and junction transistors is shown in
figure 9. r., rN, and rT, are dynamic resistances
which can be associated with the emitter, base
and collector regions of the transistor. The
current generator aI., represents the transport
of charge from emitter to collector. Typical
values of the equivalent circuit are shown in
figure 9.
Transistor
Configurations
There are three basic transistor configurations: grounded
base connection, grounded
emitter connection, and grounded collector
connection. These correspond roughly to
grounded grid, grounded cathode, and grounded plate circuits in vacuum tube terminology
(figure 10) .
The grounded base circuit has a low input
impedance and high output impedance, and no
phase reversal of signal from input to output
circuit. The grounded emitter circuit has a
higher input impedance and a lower output
impedance than the grounded base circuit, and
a reversal of phase between the input and output signal occurs. This circuit usually provides
maximum voltage gain from a transistor. The
grounded collector circuit has relatively high
input impedance, low output impedance, and
no phase reversal of signal from input to output circuit. Power and voltage gain are both
low.
Figure 11 illustrates some practical vacuum
tube circuits, as applied to transistors.
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Transistors and Semi -Conductors
96
GROUNDED EMITTER
CONNECTION
GROUNDED BASE
CONNECTION
THE RADIO
GROUNDED COLLECTOR
CONNECTION
Figure 10
COMPARISON OF BASIC VACUUM TUBE AND TRANSISTOR CONFIGURATIONS
5 -5
Transistor Circuitry
To establish the correct operating parameters
of the transistor, a bias voltage must be established between the emitter and the base. Since
transistors are temperature sensitive devices,
and since some variation in characteristics usually exists between transistors of a given type,
attention must be given to the bias system to
FLIP -FLOP COUNTER
R. F.
overcome these difficulties. The simple self -bias
system is shown in figure 12A. The base is
simply connected to the power supply through
a large resistance which supplies a fixed value
of base current to the transistor. This bias
system is extremely sensitive to the current
transfer ratio of the transistor, and must be adjusted for optimum results with each transistor.
When the supply voltage is fairly high and
OSCILLATOR
ONE -STAGE RECEIVER
RFC
CRYSTAL OSCILLATOR
BLOCKING OSCILLATOR
DIRECT -COUPLED AMPLIFIER
Figure 11
TYPICAL TRANSISTOR CIRCUITS
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AUDIO AMPLIF ER
HANDBOOK
Transistor Circuitry
97
-E
E
BIAS
BIAS
LOAD
RESISTOR
RESISTOR
RESISTOR
LOAD
RESISTOR
LOAD
RESISTOR
R2= lo Re
Re = soo- +000
2
SO
+
e
O
n
Li
(REVERSE POLARITY
FOR NPN TRANSISTOR)
O
Figure 12
BIAS CONFIGURATIONS FOR TRANSISTORS.
The voltage divider system of C is recommended for general transistor use. Ratio of
establishes base bias, and emitter bias is provided by voltage drop across Re.
Battery Polarity is reversed for N -P -N transistors.
wide variations in ambient temperature do not
occur, the bias system of figure 12B may be
used, with the bias resistor connected from
base to collector. When the collector voltage
is high, the base current is increased, moving
the operating point of the transistor down the
load line. If the collector voltage is low. the
operating point moves upwards along the load
line, thus providing automatic control of the
base bias voltage. This circuit is sensitive to
changes in ambient temperature, and may permit transistor failure when the transistor is
operated near maximum dissipation ratings.
A better bias system is shown in figure 12C,
where the base bias is obtained from a voltage
divider, (R1, R2 ), and the emitter is forward
biased. To prevent signal degeneration, the
emitter bias resistor is bypassed with a large
capacitance. A high degree of circuit stability
is provided by this form of bias, providing the
emitter capacitance is of the order of 50 t+fd.
for audio frequency applications.
Audio Circuitry
A simple voltage amplifier
is shown in figure 13. Distabilization is employed in the
rect current
enitter circuit. Operating parameters for the
R1
/R:
amplifier are given in the drawing. In this case,
the input impedance of the amplifier is quite
low. When used with a high impedance driving source such as a crystal microphone a step
down input transformer should be employed
as shown in figure 13B. The grounded collector circuit of figure 13C provides a high input
impedance and a low output impedance, much
as in the manner of a vacuum tube cathode
follower.
The circuit of a two stage resistance coupled
amplifier is shown in figure 14A. The input
impedance is approximately 1100 ohms. Feedback may be placed around this amplifier from
the emitter of the second stage to the base of
the first stage, as shown in figure 14B. A
direct coupled version of the r -c amplifier is
shown in figure 14C. The input impedance is
of the order of 15,000 ohms, and an overall
voltage gain of 80 may be obtained with a
supply potential of 12 volts.
It is possible to employ N -P -N and P -N -P
transistors in complementary symmetry circuits
which have no equivalent in vacuum tube design. Figure 15A illustrates such a circuit. A
symmetrical push -pull circuit is shown in
-t2V
VOLTAGE GAIN = 80
INPUT IMPEDANCE L 1200
VOLTAGE GAI N
P -N
.
0 97
INPUT IMPEDANCE 'L 300 /l
11
Figure 13
VOLTAGE AMPLIFIERS
-P TRANSISTOR
11
A resistance coupled amplifier employing an inexpensive CK -722 transistor is shown in A. For use with
a high impedance crystal microphone, a step -down transformer matches the low input impedance of the
transistor, as shown in B. The grounded collector configuration of C provides an input impedance of
about 300,000 ohms.
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98
THE RADIO
Transistors and Semi -Conductors
4.7N
6A/N'L
IOUF
R2
Figure 14
TWO STAGE TRANSISTOR AUDIO AMPLIFIERS
The feedback loop of B may be added to the r -c amplifier to reduce distortion, or to control the
audio response. A direct coupled amplifier is shown in C.
figure 15B. This circuit may be used to directly drive a high impedance loudspeaker,
eliminating the output transformer. A direct
coupled three stage amplifier having a gain
figure of 80 db is shown in figure 15C.
The transistor may also be used as a class
A power amplifier, as shown in figure 16A.
Commercial transistors are available that will
provide five or six watts of audio power when
operating from a 12 volt supply. The smaller
units provide power levels of a few milli watts. The correct operating point is chosen so
that the output signal can swing equally in the
positive and negative directions, as shown in
the collector curves of figure 16B.
The proper primary impedance of the output transformer depends upon the amount of
power to be delivered to the load:
RP
E:
2R.
The collector current bias is:
Ic-=
213"
E,
In a class A output stage, the maximum a -c
power output obtainable is limited to 0.5 the
allowable dissipation of the transistor. The
product I, E, determines the maximum collector
dissipation, and a plot of these values is shown
in figure 16B. The load line should always lie
under the dissipation curve, and should encompass the maximum possible area between the
axes of the graph for maximum output condition. In general, the load line is tangent to the
dissipation curve and passes through the supply
voltage point at zero collector current. The d -c
operating point is thus approximately one -half
the supply voltage.
The circuit of a typical push -pull class B
transistor amplifier is shown in figure 17A.
Push -pull operation is desirable for transistor
operation, since the even -order harmonics are
largely eliminated. This permits transistors to
be driven into high collector current regions
without distortion normally caused by non linearity of the collector. Cross -over distortion
is reduced to a minimum by providing a slight
forward base bias in addition to the normal
emitter bias. The base bias is usually less than
0.5 volt in most cases. Excessive base bias will
boost the quiescent collector current and thereby lower the overall efficiency of the stage.
2N78
NPN
PNP
NPN
2N78
NPN
2N77
PNP
+E
só_11
SPEAKER
O
O
Figure 15
COMPLEMENTARY SYMMETRY AMPLIFIERS.
N -P -N and P -N -P transistors may be combined in circuits which have no equivalent in vacuum tube
design. Direct coupling between cascaded stages using a single power supply source may be employed, as
in C. Impedance of power supply should be extremely low.
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HANDBOOK
Transistor Circuitry
99
2N 187A
I MAX
MAXIMUM COLLECTOR
DISSIPATION (IC X EC)
Figure 16
TYPICAL CLASS -A
AUDIO POWER
OPERATING POINT
TRANSISTOR CIRCUIT.
The
correct operating point
is
:hosen so that output signal can
swing equally in a positive or
negative direction, without ex:ceding maximum collector dis-
The operating point of the class B amplifier is set on the I. =O axis at the point where
the collector voltage equals the supply voltage.
The collector to collector impedance of the
output transformer is:
Rc-.
=
2Er'
Po
In the class B circuit, the maximum a -c
power input is approximately equal to five
times the allowable collector dissipation of
each transistor. Power transistors, such as the
2N301 have collector dissipation ratings of
5.5 watts and operate with class B efficiency
of about 67%. To achieve this level of operation the heavy duty transistor relies upon efficient heat transfer from the transistor case
to the chassis, using the large thermal capacity
of the chassis as a heat sink. An infinite heat
sink may be approximated by mounting the
transistor in the center of a 6" x 6" copper or
aluminum sheet. This area may be part of a
'arger chassis.
The collector of most power transistors is
electrically connected to the case. For applications where the collector is not grounded a
thin sheet of mica may be used between the
case of the transistor and the chassis.
Power transistors such as the Philco T -1041
may be used in the common collector class B
a.7N
configuration (figure 17C) to obtain high
power output at very low distortions comparable with those found in quality vacuum tube
circuits having heavy overall feedback. In addition, the transistor may be directly bolted to
the chassis, assuming a negative grounded
power supply Power output is of the order of
10 watts, with about 0.5% total distortion.
Circuitry
Transistors may be used for
radio frequency work provided
the alpha cutoff frequency of the units is
sufficiently higher than the operating frequency. Shown in figure 18A is a typical i -f
amplifier employing an N -P -N transistor. The
collector current is determined by a voltage
divider on the base circuit and by a bias resistor in the emitter leg. Input and output are
coupled by means of tuned i -f transformers.
Bypass capacitors are placed across the bias
resistors to prevent signal frequency degeneration. The base is connected to a low impedance untuned winding of the input transformer, and the collector is connected to a tap
on the output transformer to provide proper
matching, and also to make the performance of
the stage relatively independent of variations
between transistors of the same type. With a
rate -grown N -P -N transistor such as the G.E.
2N293, it is unnecessary to use neutralization
to obtain circuit stability. When P -N -P alloy
R -F
2N225
V
T-104_1
12V.
ZP-soonc.T.
ZS'3000CT.
2Ec
Ec
COLLECTOR VOLTAGE
sipation.
z
ZS=
LOAD
BOO
n
LINE
200 MW
NO SIGNAL
OPERATING
POINT
J
2N109
-T
R1
COLLECTOR VOLTAGE
EC
SO
ADJUST Ri FOR
V. BASE BIAS
O.4
1
X
ICC
ICC
í13v.
- 0.3 AMP.
(NAB.)',.35A.
PO'
10 WATTS
Figure 17
CLASS -B AUDIO AMPLIFIER CIRCUITRY.
C
permits
the
The common collector circuit of
transistor to be bolted directly to the chassis for efficient
heat transfer from the transistor case to the chassis.
100
Transistors and Semi- Conductors
2N293
N
THE RADIO
2N135
PN
p
TO
e
I
OUT.
MIXER
OR
P.
OUT.
TO
MIXER
OR
CONVERTER
CONVERTER
T N
r9v
Figure 18
TRANSISTORIZED I -F AMPLIFIERS.
Typical
transistor must be neutralized because of high collector capacitance.
grown N -P -N transistor does not usually require external neutralizing circuit.
P -N -P
Rate
1N64
Figure 19
AUTOMATIC VOLUME
CONTROL CIRCUIT
ro
MIXER
OR
CONVERTER
FOR TRANSISTORIZED
I -F
AMPLIFIER.
transistors are used, it is necessary to neutralize the circuit to obtain stability (figure 18B).
The gain of a transistor i -f amplifier will
decrease as the emitter current is decreased.
This transistor property can be used to control
the gain of an i -f amplifier so that weak and
strong signals will produce the same audio
output. A typical i -f strip incorporating this
automatic volume control action is shown in
figure 19.
R -f transistors may be used as mixers or
autodyne converters much in the same manner
as vacuum tubes The autodyne circuit is shown
in figure 20. Transformer T, feeds back a
signal from the collector to the emitter causing oscillation. Capacitor C, tunes thé oscillator
circuit to a frequency 455 kc. higher than that
of the incoming signal. The local oscillator
signal is inductively coupled into the emitter
circuit of the transistor. The incoming signal
is resonated in T_ and coupled via a low impedance winding to the base circuit. Notice
that the base is biased by a voltage divider
circuit much the same as is used in audio frequency operation. The two signals are mixed
in this stage and the desired beat frequency of
455 kc. is selected by i -f transformer T1 and
passed to the next stage. Collector currents of
0.6 ma. to 0.8 ma. are common, and the local
oscillator injection voltage at the emitter is in
the range of 0.15 to 0.25 volts, r.m.s.
A complete receiver "front end" capable of
operation up to 23 Mc. is shown in figure 21.
The RCA 2N247 drift transistor is used for
the r -f amplifier (TRI ), mixer (TR2), and
high frequency oscillator (TR3) The 2N247
incorporates an interlead shield, cutting the
interlead capacitance to .003 q fd. If proper
shielding is employed between the tuned circuits of the r -f stage, no neutralization of the
stage is required. The complete assembly obtains power from a 9 -volt transistor battery.
Note that input and output circuits of the transistors are tapped at low impedance points on
the r -f coils to achieve proper impedance match.
.
Figure 20
THE AUTODYNE CONVERTER CIRCUIT
USING A 2N168A AS A MIXER.
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HANDBOOK
RF
Transistor Circuitry
101
Figure 21
AMPLIFIER, MIXER,
AND OSCILLATOR
STAGES FOR
TRANSISTORIZED
HIGH FREQUENCY
RECEIVER. THE RCA
2N247 DRIFT
TRANSISTOR IS
CAPABLE OF
EFFICIENT OPERATION
UP TO 23 Mc.
i
L
Sufficient coupling of the proper
input and output
circuits of the transistor will permit oscillation up to and slightly above the
alpha cutoff frequency. Various forms of transistor oscillators are shown in figure 22. A
simple grounded emitter Hartley oscillator having positive feedback between the base and the
collector (22A) is compared to a grounded
base Hartley oscillator (22B) . In each case
the resonant tank circuit is common to the input and output circuits of the transistor. Self bias of the transistor is employed in both these
circuits A more sophisticated oscillator employing a 2N247 transistor and utilizing a
voltage divider -type bias system (figure 22C)
is capable of operation up to 50 Mc. or so.
The tuned circuit is placed in the collector,
with a small emitter -collector capacitor providing feedback to the emitter electrode.
A P -N -P and an N -P -N transistor may be
combined to form a complementary Hartley
oscillator of high stability ( figure 23). The
collector of the P -N -P transistor is directly
Transistor
Oscillators
phase between
coupled to the base of the N -P -N transistor,
and the emitter of the N -P -N transistor furnishes the correct phase reversal to sustain oscillation. Heavy feedback is maintained between the emitter of the P -N -P transistor and
the collector of N -P -N transistor. The degree
of feedback is controlled by R1. The emitter
resistor of the second transistor is placed at the
+9V.
PwP
2N247
1N81
P -N
1N81
ion
NPN
2N78
Figure 23
COMPLEMENTARY HARTLEY
OSCILLATOR
-P and N -P -N transistors
bility
form high sta-
oscillator. Feedback between P -N -P
emitter and N -P -N collector is controlled by
R,. 1N81 diodes are used as amplitude limiters. Frequency of oscillation is determined
by L, C, -C :.
RFC
RFC
Figure 22
TYPICAL TRANSISTOR OSCILLATOR CIRCUITS
A- Grounded
Emitter Hartley
Grounded Base Hartley
C -2N247 Oscillator Suitable for
B-
SO
Mc. operation.
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102
THE RADIO
Transistors and Semi -Conductors
2N33
POiNT-CONTA.T
TRANS.STON
POINT -CONTACT TRANSISTOR
LE
CI.ARGN
PER OO
RFC
:E
V.
Figure 25
RELAXATION OSCILLATOR USING
POINT- CONTACT OR SURFACE
Figure 24
NEGATIVE RESISTANCE OF
POINT -CONTACT TRANSISTOR
PERMITS HIGH FREQUENCY
OSCILLATION (50 Mc) WITHOUT
WITHOUT NECESSITY OF
EXTERNAL FEEDBACK PATH.
BARRIER TRANSISTORS.
Relaxation
Oscillators
Transistors have almost unlimited use in relaxation acid R -C oscillator service. The negative re-
sistance characteristic of the point contact transistor make it well suited to such application.
Surface barrier transistors are also widely used
in this service, as they have the highest alpha
cutoff frequency among the group of -alphaless-than-unity- transistors. Relaxation oscillators used for high speed counting require transistors capable of operation at repetition rates
of 5 Mc. to 10 Mc.
center of the oscillator coil to eliminate loading of the tuned circuit.
Two germanium diodes are employed as
amplitude limiters, further stabilizing amplifier operation. Because of the low circuit impedances, it is permissible to use extremely
high -C in the oscillator tank circuit, effectively
limiting oscillator temperature stability to variations in the tank inductance.
The point- contact transistor exhibits negative input and output resistances over part of
its operaing range, due to its unique ability
to multiply the input current. This characteristic affords the use of oscillator circuitry having no external feedback paths ( figure 24).
A high impedance resonant circuit in the base
lead produces circuit instability and oscillation
at the resonant frequency of the L -C circuit.
Positive emitter bias is used to insure thermal
circuit stability.
A simple emitter controlled relaxation oscillator is shown in figure 25, together with
its operating characteristic. The emitter of the
transistor is biased to cutoff at the start of the
cycle (point I) The charge on the emitter capacitor slowly leaks to ground through the
emitter resistor, R1. Discharge time is determined by the time constant of RICI. When the
emitter voltage drops sufficiently low to permit
the transistor to reach the negative resistance
region (point 2) the emitter and collector resistances drop to a low value, and the collector
.
*E
e.
PO5ITiwE
TRi.,,ER
PJLSE
EPUL 5E
M
OUT
NPN
-
1/111
NP
P N P
lON
Figure 26
TRANSISTORIZED BLOCKING OSCILLATOR (A) AND ECCLES -JORDAN
BI- STABLE MULTIVIBRATOR (B).
High -alpha transistors must be employed in counting circuits to reduce effects of
storage time caused by transit lag in transistor base.
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HANDBOOK
Transistor Circuitry
103
2n
PHONE`,
C1- 123LLF, .W.
M /LLER
# 2110
C2-
791/11F, PART OF C I
L1 - "LOOPSTIC K. COIL, J.W MILLER
La- OSCILLATOR
COIL, J
W.
I
Figure 28
SCHEMATIC, TRANSISTORIZED BROADCAST BAND (S00
DYNE RECEIVER.
"
L OOP ST ICI
COIL
Figure 27
"WRIST RADIO" CAN BE MADE
WITH LOOPSTICK, DIODE, AND
INEXPENSIVE CK -722 TRANSISTOR.
A TWENTY FOOT ANTENNA WIRE
WILL PROVIDE GOOD RECEPTION
IN STRONG SIGNAL AREAS.
current is limited only by the collector resistor,
R. The collector current is abruptly reduced
by the charging action of the emitter capacitor
CI (point 3), bringing the circuit back to the
original operating point. The "spike" of collector current is produced during the charging
period of C. The duration of the pulse and the
pulse repetition frequency (p.r.f.) are controlled by the values of C, R1, R_, and R.
Transistors may also be used as blocking
oscillators (figure 26A) . The oscillator may
be synchronized by coupling the locking signal
to the base circuit of the transistor. An oscillator of this type may be used to drive a flip flop circuit as a counter. An Eccles -Jordan
bi- stable flip -flop circuit employing surface barrier transistors may be driven between "off"
and "on" positions by an exciting pulse as
shown in figure 26B. The first pulse drives
the "on" transistor into saturation. This transistor remains in a highly conductive state until
the second exciting pulse arrives. The transistor does not immediately return to the cut -off
-
#2003
MILLER 4 2002
TI -4SS RC. I.F. TRANSFORMER,
T2 -455 PI C. F. TRANSFORMER,
J.W
MILLER2031
J.W. MILLER
2032
1600 KC.) SUPERHETERO-
state, since a time lapse occurs before the output waveform starts to decrease. This storage
time is caused by the transit lag of the minority
carriers in the base of the transistor. Proper circuit design and the use of high -alpha transistors can reduce the effects of storage time to a
minimum. Driving pulses may be coupled to
the multivibrator through steering diodes as
shown in the illustration.
5 -6
Transistor Circuits
With the introduction of the dollar transistor, many interesting and unusual experiments and circuits may be built up by the beginner in the transistor field. One of the most
interesting is the "wrist watch" receiver, illustrated in figure 27. A diode and a transistor
amplifier form a miniature broadcast receiver,
which may be built in a small box and carried
on the person. A single 1.5 -volt penlite cell
provides power for the transistor, and a short
length of antenna wire will suffice in the vicinity of a local broadcasting station.
A transistorized superhetrodyne for broadcast reception is shown in figure 28. No antenna is required, as a ferrite "loop-stick" is
used for the r -f input circuit of the 2N136
mixer transistor. A miniature magnetic "hearing aid" type earphone may be employed with
this receiver.
A simple phonograph
amplifier designed
for use with a high impedance crystal pickup
is shown in figure 29. Two stages of amplification using 2N109 transistors are used to
drive two 2N109 transistors in a class B con-
figuration. Approximately 200 milliwatts of
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104
Transistors and Semi -Conductors
2N109
2N1O9
220N
CRYSTAL
PICKUP
f0
0
i5M
2UF
2N109
0
27M
K
Figure 29
HIGH GAIN, LOW DISTORTION AUDIO AMPLIFIER, SUITABLE FOR USE
WITH A CRYSTAL PICKUP. POWER OUTPUT IS 250 MILLIWATTS.
power may be obtained with a battery supply
of 12 volts. Peak current drain under maximum signal conditions is 40 ma.
Shown in figure 30 is an inexpensive and
compact 25 watt transistorized modulator suitable for mobile use with an automobile having
a 12 volt ignition system. This unit may be
used to modulate a 6146 r.f. amplifier stage
running at 400 volts and 125 milliamperes
plate power input. The two DS -501 power
transistors (Delco) are mounted on a heat
sink made of a 6" x 6" x 1/8" aluminum plate.
The components are mounted on the sink
which serves as the chassis. Output transformer
T1
-I50
T., consists of a 6.3 volt filament transformer
with the ends of the low voltage winding connected to the collectors of the output transistors. Resting modulator current is about 0.7
amperes, rising to nearly 2 amperes on full
modulation peaks.
The modulator should be positioned so that
motor heat and warm air is deflected from the
unit, or the efficiency of the aluminum heat
sink will be impaired and damage to the output transistors may result. A good location for
the unit is under the dash against the firewall.
Microphone gain may be adjusted by changing the value of the 100 ohm, 2 watt series
resistor.
Figure 30.
TRANSISTORIZED 25 WATT MOBILE MODULATOR.
ohm primary, 490 ohm secondary (center tap on primary not used), Thordarson
TR -S.
T2-400 ohm primary, 4 and 16 ohm secondary. Stancor TA -41.
T3-6.3 volt center tap, 3a. (See text.)
Note-Output transistors are insulated from heat-sink by Delco
(s: 1221264).
insulators
mica
R.F.
LOAD
NOTE:
C /ACV /T RETURNS ro
BATTERY TERMINAL.
CONTROL
-Nor-
CONTROL
CIRCUIT
+12.e
V.
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-1z.e
V.
Zener Diodes
5 -7
Zener Diodes
The Zener Diode is a semiconductor device
that can be used as a constant voltage reference, or as a control element. Zener diodes
are available in ratings to 50 watts, with zener
voltages of approximately 4 volts to 200 volts.
The zener diode has electrical characteristics that are derived from a rectifying junction which operates at a reverse bias condition
not normally used. The zener knee (figure 31)
and constant voltage plateau are obtained
when this rectifying junction is back -biased
above the junction breakdown voltage. The
break from non -conductance to conductance is
very sharp. At applied voltages greater than
the breakdown point, the voltage drop across
the diode junction becomes essentially constant for a relatively wide range of currents.
This is the zener control region.
Thermal dissipation is obtained by mounting the zener diode to a heat sink composed
of a large area of metal having free access
to ambient air.
-
ZENER KNEE
MA.
CONSTANT
VOLTAGE
Mi
M
_
CURRENT
1.S
1S
2
VP (VOLTS)
u
REVERSe
CHARACTERISTIC
S=
{M
AM.
N
FIGURE
31
BETWEEN .ZENER KNEE- AND POINT OF MAXIMUM ZENER CUR RENT, THE ZENER VOLTAGE IS ESSENTIALLY CONSTANT AT SOVOLTS.
DIODE
2 VOLT
REG.
VOLT.
FIGURE 32
-ZENER DIODE FUNCTIONS AS
VOLTAGE REGULATOR OVER
RANGE OF CONSTANT VOLTAGE
PLATEAU.
B -TWO ZENER DIODES OF DIFFER
OS
1.0
MAX.ZENER
10 v
1LOUT
MM
iM=
MlAT 5
The zener diode may be employed as a shunt regulator
(figure 32A) in the manner
of a typical "VR- tube." Two zener diodes may
be employed in the circuit of figure 32B to
supply very low values of regulated voltage.
Two opposed zener diodes can be used to provide a.c. clipping of both halves of the cycle
(figure 32C) . Zener diodes may also be used
to protect meter movements as they provide a
very low resistance shunt across the movement when the applied voltage exceeds a
certain critical level.
IN ti
AMMO
REVERSE VOLTAGE
10
30
20
Applications
A
2 10
CHARACTERISTIC
Zener Diode
UNREG.
VOLT
qM/iI
MMMMM
,.
MUM=
105
-
ENT VOLTAGE CAN PROVIDE
SMALL REGULATED VOLTAGE.
C- OPPOSED ZENER DIODES CLIP
BOTH HALVES OF CYCLE OF A.C.
WAVE.
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CHAPTER SIX
Vacuum Tube Amplifiers
6 -1
Vacuum Tube Parameters
Electrode Potentials
Symbols for
Vacuum -Tube
Parameters
E«
ep
eg
Ep
Eg
grid- screen
conversion
`Bp
Cpi,
Cm
Cou,
I,
fpm
ipmax
igmaz
Ip
Ig
factor
tube)
grid
grid voltage
cutoff
average
grid current
fundamental
maximum
maximum
grid current
Other Symbols
grid- cathode
grid
Pi
plate- cathode
input
output
grid
minimum
maximum
-- average plate current
-peak
plate
-instantaneous plate current
instantaneous
-- static
plate current
static
- plate
P.-plate
-- plate dissipation plus bias losses)
lb
Intere!ectrode Capacitances
C¡gk
-- instantaneous
plate potential
instantaneous
potential
instantaneous plate voltage
-positive instantaneous
voltage
static plate voltage
---static
bias
Electrode Currents
factor
mu
-
eco
Gm
Gc
grid supply voltage (a negative
epmin
egmp
Tube Constants
--transconductance
plate resistance
/tu transconductance(mixer
-- -plate capacitance
capacitance
capacitance
--- capacitance
(tetrode
capacitance (tetrode
-d-c
peak grid excitation voltage (1/2 total
peak -to -peak grid swing)
Epm -peak plate voltage (! i total peak -to -peak
plate swing)
Egm
involving vacuum -tube parameters, the following symbols
will be used throughout this book:
R,
plate supply voltage (a positive
quantity)
quantity)
As an assistance in simplify ing and shortening expressions
ft- amplification
-d-c
Ebb
The ability of the control grid of a vacuum
tube to control large amounts of plate power
with a small amount of grid energy allows the
vacuum tube to be used as an amplifier. It is
this ability of vacuum tube s to amplify an
extremely small amount of energy up to almost
any level without change in anything except
amplitude which makes the vacuum tube such
an extremely valuable adjunct to modern electronics and communication.
or pentode)
or pentode)
Pp
Pd
power input
power output
grid driving power (grid
106
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current
grid current
Classes of Amplifiers
ST
-F-_,
r--
r--1
17-T-
Cw ---
Ccv:7:
-
I
.7.-.7.
CcKi_
:=.
L
Clan
CiN
:t:
-,Cour
I
__ _J
1
PENTODE OR TETRODE
TRIODE
Figure
STATIC
INTERELECTRODE
TANCES WITHIN A TRIODE,
OR TETRODE
1
CAPACIPENTODE,
107
is the grid -to -plate capacitance, and A is
the stage gain. This expression assumes that
the vacuum tube is operating into a resistive
load such as would be the case with an audio
stage working into a resistance plate load in
the middle audio range.
The more complete expression for the input
admittance (vector sum of capacitance and
resistance) of an amplifier operating into any
type of plate load is as follows:
Cgp
Input capacitance = Cgk
+ (1 + A
cos
(9) Cgp
Input resistance
135-grid dissipation
Np
9p
9g
R1
Z1
--load
-
plate efficiency(expressed as a decimal)
one -half angle of plate current flow
one -half angle of grid current flow
resistance
load impedance
A
The relationships between certain of the electrode potentials
and currents within a vacuum
tube are reasonably constant under specified
alone
angle of the plate load impedance, positive for inductive
loads, negative for capacitive
Vacuum-Tube
conditions of operation. These relationships
are called vacuum -tube constants and are
listed in the data published by the manufacturers of vacuum tubes. The defining equations
for the basic vacuum -tube constants are given
in Chapter Four.
Interelectrode
The values of interelectrode
Capacitances and capacitance published in
Miller Effect
vacuum -tube tables are the
static values measured, in
the case of triodes for example, as shown in
figure 1. The static capacitances are simply
as shown in the drawing, but when a tube is
operating as amplifier there is another consideration known as Miller Effect which causes
the dynamic input capacitance to be different
from the static value. The output capacitance
of an amplifier is essentially the same as the
static value given in the published tube tables.
The grid -to-plate capacitance is also the same
as the published static value, but since the
Cgp acts as a small capacitance coupling energy back from the pl ate circuit to the grid
circuit, the dynamic input capacitance is equal
ro the static value plus an amount (frequently
much greater in the case of a triode) determined by the gain of the stage, the plate load
impedance, and the Cgp feedback capacitance.
The total value for an audio amplifier stage
can be expressed in the following equation:
(dynamic) _
(
static)
+
(A
+ 1) Cgp
where Co is the grid -to- cathode capacitance,
O
= phase
O
Constants
sin
Where: Cgk = grid -to- cathode capacitance
Cgp = grid -to-plate capacitance
A
= voltage amplification of the tube
It can be seen from the above that if the
plate load impedance of the stage is capacitive or inductive, there will be a resistive component in the input admittance of the stage.
The resistive component of the input admittance will be positive (tending to load the
circuit feeding the grid) if the load impedance
of the plate is capacitive, or it will be negative
(tending to make the stage oscillate) if the
load impedance of the plate is inductive.
Neutralization
of Interelectrode
Capacitance
Neutralization of the effects
of interelectrode capacitance
is employed most frequently
in the case of radio frequency power amplifiers. Before the introduction of the tetrode and pentode tube, triodes
were employed as neutralized Class A amplifiers in receivers. This practice has been
largely superseded in the present state of the
art through the use of tetrode and pentode
tubes in which the Cge or feedback capacitance has been reduced to such a low value
that neutralization of its effects is not necessary to prevent oscillation and instability.
6 -2
Classes and Types of
Vacuum -Tube Amplifiers
Vacuum -tube amplifiers are grouped into
various classes and sub -classes according to
the type of work they are intended to perform.
The difference between the various classes is
determined primarily by the value of average
grid bias employed and the maximum value of
www.americanradiohistory.com
108
the
grid.
Vacuum
Tube
Amplifiers
THE
exciting signal to he impressed upon the
A Class A amplifier is an amplifier
biased and supplied with excitation
Class A
Amplifier
of such amplitude that plate current flows continuously (360° of the exciting
voltage waveshape) and grid current does not
flow at any time. Such an amplifier is normally
operated in the center of the grid- voltage
plate- current transfer characteristic and gives
an output waveshape which is a substantial
replica of the input waveshape.
Class Al
Amplifier
This is another term applied to the
Class A amplifier in which grid
current does not flow over any
portion of the input wave cycle.
This is
a Class A amplifier operated under such conditions that the
grid is driven positive over a portion of the input voltage cycle, but plate current still flows over the entire cycle.
Class A2
Amplifier
Class AB1 This is an amplifier operated under
Amplifier
such conditions of grid bias and
exciting voltage that plate current
flows for more than one-half the input voltage
cycle but for less than the complete cycle. In
other words the operating angle of plate current flow is appreciably greater than 180° but
less than
360°. The suffix 1 indicates that grid
current does not flow over any portion of the
input cycle.
Class AB2 amplifier is operated
under essentially the same conditions of grid bias as the Class
AB t amplifier mentioned above, but the exciting voltage is of such amplitude that grid current flows over an appreciable portion of the
input wave cycle.
Class AB2
A
Amplifier
amplifier is biased sub stantially to cutoff of plate current
(without exciting voltage) so that
plate current flows essentially over one -half
Class
B
A Class B
Amplifier
the input voltage cycle. The operating angle
of plate current flow is essentially 180 °. The
Class B amplifier is almost always excited
to such an extent that grid current flows.
Class C amplifier is biased to a
Amplifier value greater than the value required for plate current cutoff and
is excited with a signal of such amplitude
that grid current flows over an appreciable
period of the input voltage waveshape. The
angle of plate current flow in a Class C amplifier is appreciably less than 180 °, or in
other words, plate current flows appreciably
Class
RADIO
C
A
Figure 2
TYPES OF BIAS SYSTEMS
A
B
-
C -
Grid bias
Cathode bias
Grid leak bias
less than one -half the time. Actually, the conventional operating conditions for a Class C
amplifier are such that plate current flows for
120° to 150° of the exciting voltage waveshape.
There are three general types of
amplifier circuits in use. These
types are classified on the basis
of the return for the input and output circuits.
Conventional amplifiers are called cathode return amplifiers since the cathode is effectively
grounded and acts as the common return for
both the input and output circuits. The second
type is known as a plate return amplifier or
cathode follower since the plate circuit is effectively at ground for the input and output
signal voltages and the output voltage or
power is taken between cathode and plate. The
third type is called a grid -return or grounded grid amplifier since the grid is effectively at
ground potential for input and output signals
and output is taken between grid and plate.
Types of
Amplifiers
6 -3
Biasing Methods
The difference of potential between grid and
cathode is called the grid bias of a vacuum
tube. There are three general methods of
providing this bias voltage. In each of these
methods the purpose is to establish the grid
at a potential with respect to the cathode
which will place the tube in the desired operating condition as determined by its charac-
teristics.
Grid bias may be obtained from a source of
voltage especially provided for this purpose,
as a battery or other d -c power supply. This
method is illustrated in figure 2A, and is
known as fixed bias.
A second biasing method is illustrated in
figure 2B which utilizes a cathode resistor
across which an IR drop is developed as a
result of plate current flowing through it. The
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Amplifier
HANDBOOK
cathode of the tube is held at a positive potential with respect to ground by the amount of
the IR drop because the grid is at ground potential. Since the biasing voltage depends
upon the flow of plate current the tube cannot
be held in a cutoff condition by means of the
cat bode bias voltage developed across the
cathode resistor. The value of this resistor is
determined by the bias required and the plate
current which flows at this value of bias, as
found from the tube characteristic curves.
A capacitor is shunted across the bias resistor
to provide a low impedance path to ground for
the a -c component of the plate current which
results from an a-c input signal on the grid.
The third method of providing a biasing
voltage is shown in figure 2C, and is called
grid -leak bias. During the portion of the input
cycle which causes the grid to be positive
with respect to the cathode, grid current flows
from cathode to grid, charging capacitor C,.
When the grid draws current, the grid -to- cathode
resistance of the tube drops from an infinite
value to a very low value, on the order of
1,000 ohms or so, making the charging time
constant of the capacitor very short. This enables Cs to charge up to essentially the full
value of the positive input voltage and results
in the grid (which is connected to the low potential plate of the capacitor) being held essentially at ground potential. During the negative swing of the input signal no grid current
flows and the discharge path of Cg is through
the grid resistance which has a value of
500,000 ohms or so. The discharge time constant for C5 is, therefore, very long in comparison to the period of the input signal and
only a small part of the charge on C5 is lost.
Thus, the bias voltage developed by the discharge of Cs is substantially constant and the
grid is not permitted to follow the positive
portions of the input signal.
Distortion in Amplifiers
6 -4
There are three main types of distortion that
may occur in amplifiers: frequency distortion,
phase distortion and amplitude distortion.
distortion may occur
when some frequency components
of a signal are amplified more than
Frequency distortion occurs at low
Frequency
Distortion
Distortiob
109
OUTPUT
SIGNAL
Figure
3
Illustration of the effect of phase distortion
on input wave containing o third harmonic
signal
two stage amplifier. Although the amplitudes
of both components are amplified by identical
ratios, the output waveshape is considerably
different from the input signal because the
phase of the third harmonic signal has been
shifted with respect to the fundamental signal.
This phase shift is known as phase distortion,
and is caused principally by the coupling circuits between the stages of the amplifier.
Most coupling circuits shift the phase of a
sine wave, but this has no effect on the shape
of the output wave. However, when a complex
wave is passed through the same coupling
circuit, each component frequency of the waveshape may be shifted in phase by a different
amount so that the output wave is not a faithful reproduction of the input waveshape.
a
Amplitude
Distortion
If
a signal is passed through a vac uum tube that is operating on any
non -linear part of its characteristic,
amplitude distortion will occur. In such a region, a change in grid voltage does not result
in a change in plate current which is directly
proportional to the change in grid voltage. For
example, if an amplifies is excited with a signal that overdrives the tubes, the resultant
signal is distorted in amplitude, since the
tubes operate over a non -linear portion of their
characteristic.
Frequency
others.
frequencies if coupling capacitors between
stages are too small, or may occur at high frequencies as a result of the shunting effects of
the distributed capacities in the circuit.
Phase
Distortion
input signal con sisting of a fundamental and a
third harmonic is passed through
In
figure
3
an
6 -5
Resistance Capacitance Coupled
Audio -Frequency Amplifiers
Present practice in the design of audio-frequency voltage amplifiers is almost exclusively
to use resistance -capacitance coupling between the low -level stages. Both triodes and
www.americanradiohistory.com
1
1
Vacuum
0
Tube
A mp
l
i
fie
T H E
r s
Figure 4
CIRCUIT FOR RESISTANCE CAPACITANCE COUPLED TRIODE AMSTANDARD
PLIFIER STAGE
R A D
The voltage gain per stage of
a resistance -capacitance coupled triode amplifier can be calculated with the aid of the equivalent circuits
and expressions for the mid -frequency, high frequency, and low- frequency range given in
figure 5.
A triode R -C coupled amplifier stage is
normally operated with values of cathode resistor and plate load resistor such that the
actual voltage on the tube is approximately
one -half the d -c plate supply voltage. To
per Stage
will
be
are used; triode amplifier stages
discussed first.
R -C
Coupled
Triode Stages
4 illustrates the stand circuit for a resistance -
Figure
and
capacitance coupled amplifier
stage utilizing a triode tube with cathode bias.
In conventional audio-frequency amplifier design such stages are used at medium voltage
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levels (from 0.01 to 5 volts peak on the grid
of the tube) and use medium -p triodes such
as the 6J5 or high -p triodes such as the 6SF5
or 6SL7 -GT. Normal voltage gain for a single
stage of this type is from 10 to 70, depending
upon the tube chosen and its operating conditions. Triode tubes are normally used in the
last voltage amplifier stage of an R -C amplifier since their harmonic distortion with large
output voltage (25 to 75 volts) is less than
with a pentode tube.
Voltage Gain
pentodes
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Figure 5
Equivalent circuits and gain equations for a triode R -C coupled amplifier stage. In using these
equations, be sure to select the values of mu and RP which are proper for the static current and
voltages with which the tube will operate. These values may be obtained from curves published
in the RCA Tube Handbook RC -16.
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'7IANDBOOK
R
-C
Amplifiers
Coupled
111
such as the 6SJ7. Normal voltage gain for a
stage of this type is from 60 to 250, depending upon the tube chosen and its operating
conditions. Pentode tubes are ordinarily used
the first stage of an R -C amplifier where the
high gain which they afford is of greatest advantage and where only a small voltage output
is required from the stage.
Figure
6
CIRCUIT FOR RESISTANCE
CAPACITANCE COUPLED PENTODE AMSTANDARD
PLIFIER STAGE
assist the designer of such stages, data on
operating conditions for commonly used tubes
is published in the RCA Tube Handbook RC -16.
It is assumed, in the case of the gain equations
of figure 5, that the cathode by -pass capacitor,
Ck, has a reactance that is low with respect
to the cathode resistor at the lowest frequency
to be passed by the amplifier stage.
Coupled
Pentode Stages
R -C
6 illustrates the stand circuit for a resistance -
Figure
and
capacitance coupled pentode
amplifier stage. Cathode bias is used and the
screen voltage is supplied through a dropping
resistor from the plate voltage supply. In conventional audio -frequency amplifier design
such stages are normally used at low voltage
levels (from 0.00001 to 0.1 volts peak on the
grid of the tube) and use moderate -Gm pentodes
The voltage gain per stage of a resistance capacitance coupled pentode amplifier can be
calculated with the aid of the equivalent circuits and expressions for the mid -frequency,
high- frequency, and low- frequency range given
in figure 7.
To assist the designer of such stages, data
on operating conditions for commonly used
types of tubes is published in the RCA Tube
Handbook RC -16. It is assumed, in the case of
the gain equations of figure 7, that the cathode
by -pass capacitor, Ck, has a reactance that is
low with respect to the cathode resistor at the
lowest frequency to be passed by the stage. It
is additionally assumed that the reactance of
the screen by -pass capacitor Cd, is low with
respect to the screen dropping resistor, Rd, at
the lowest frequency to be passed by the amplifier stage.
Cascade Voltage
Amplifier Stages
When voltage amplifier stages
are operated in such a manner
that the output voltage of the
first is fed to the grid of the second, and so
forth, such stages are said to be cascaded.
The total voltage gain of cascaded amplifier
t= -GMEc
c
A
=
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RL
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Figure 7
Equivalent circuits and
gain equations for a pentode R -C coupled amplifier
stage. In using these equations be sure to select the
values of Gm and Rp which
are proper for the static
currents and voltages with
which the tube will operate. These values may be
obtained from curves published in the RCA Tube
Re
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XLL -
8
The variation of stage gain with frequency
in an r-c coupled pentode amplifier for various values of plate load resistance
WHERE
Amplifier
typical frequency response
curve for an R -C coupled audio
amplifier is shown in figure 8.
It is seen that the amplification is poor for the
extreme high and low frequencies. The reduced
gain at the low frequencies is caused by the
loss of voltage across the coupling capacitor.
In some cases, a low value of coupling capacitor is deliberately chosen to reduce the response of the stage to hum, or to attenuate
the lower voice frequencies for communication
purposes. For high fidelity work the product of
the grid resistor in ohms times the coupling
capacitor in microfarads should equal 25,000.
(ie.: 500,000 ohms x 0.05 µfd = 25,000).
The amplification of high frequencies falls
off because of the Miller effect of the subsequent stage, and the shunting effect of residual circuit capacities. Both of these effects
may be minimized by the use of a low value of
plate load resistor.
A
Response
The correct operating bias
for a high -mu triode such
as the GSL7, is fairly critical, and will be found to be highly variable
from tube to tube because of minute variations
in contact potential within the tube itself. A
satisfactory bias method is to use grid leak
bias, with a grid resistor of one to ten meg-
Grid Leak Bias
for High Mu Triodes
EQUALIZATION
XC AT fC
.
XC AT
f
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CUTOFF FRCQUENC, OF AMPLIFIER
C
fC
LL I PEAKING INDUCTOR
POR
COMPROMISE LOW PRCOUENC EQUALISATION
stages is obtained by taking the product of the
voltage gains of each of the successive stages.
R -C
S
RL
Re'
Sometimes the voltage gain of an amplifier
stage is rated in decibels. Voltage gain is
converted into decibels gain through the use
of the following expression: db = 20 log
A,
where A is the voltage gain of the stage. The
total gain of cascaded voltage amplifier stages
can be obtained by adding the number of
decibels gain in each of the cascaded stages.
0
NETWORK
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Figure 9
SIMPLE COMPENSATED VIDEO
AMPLIFIER CIRCUIT
Resistor RL in conjunction with coil LL
serves to flatten the high -frequency response
of the stage, while CB and R serve to equalize the low- frequency response of this simple video amplifier stage.
ohms connected directly between grid and
cathode of the tube. The cathode is grounded.
Grid current flows at all times, and the effective input resistance is about one -half the
resistance value of the grid leak. This circuit
is particularly well suited as a high gain
amplifier following low output devices, such
as crystal microphones, or dynamic micro-
phones.
resistance- capacity
coupled amplifier can
be designed to provide
a good frequency response for almost any
desired range. For instance, such an amplifier
can be built to provide a fairly uniform amplification for frequencies in the audio range of
about 100 to 20,000 cycles. Changes in the
values of coupling capacitors and load resistors can extend this frequency range to
cover the very wide range required for video
service. However, extension of the range can
only be obtained at the cost of reduced overall amplification. Thus the R -C method of
coupling allows good frequency response with
minimum distortion, but low amplification.
Phase distortion is less with R -C coupling
R -C Amplifier
General Characteristics
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A
HANDBOOK
Video Frequency Amplifiers
than with other types, except direct coupling.
The R -C amplifier may exhibit tendencies to
"motorboat" or oscillate if it is used with a
high impedance plate supply.
resistance -capacitance coupling is most commonly used, there are certain circuit conditions wherein coupling methods other than
resistance capacitance are more effective.
Transformer coupling, as illustrated in figure 1013, is seldom
used at the present time between
two successive single -ended stages of an
audio amplifier. There are several reasons why
resistance coupling is favored over transformer
coupling between two successive single -ended
stages. These are: (1) a transformer having
frequency characteristics comparable with a
properly designed R -C stage is very expensive;
(2) transformers, unless they are very well
shielded, will pick up inductive hum from
nearby power and filament transformers; (3)
the phase characteristics of step -up interstage
transformers are poor, making very difficult
the inclusion of a transformer of this type
within a feedback loop; and (4) transformers
are heavy.
However, there is one circuit application
where a step-up interstage transformer is of
considerable assistance to the designer; this
is the case where it is desired to obtain a
large amount of voltage to excite the grid of a
cathode follower or of a high -power Class A
amplifier from a tube operating at a moderate
plate voltage. Under these conditions it is possible to obtain apeak voltage on the secondary
of the transformer of a value somewhat greater
than the d-c plate supply voltage of the tube
supplying the primary of the transformer.
Transformer
Coupling
Video -Frequency
6 -6
Amplifiers
A video -frequency amplifier is one which
has been designed to pass frequencies from
the lower audio range (lower limit perhaps 50
cycles) to the middle r -f range (upper limit
perhaps 4 to 6 megacycles). Such amplifiers,
in addition to passing such an extremely wide
frequency range, must be capable of amplifying this range with a minimum of amplitude,
phase, and frequency distortion. Video amplifiers are commonly used in television, pulse
communication, and radar work.
Tubes used in video amplifiers must have
high ratio of Gm to capacitance if a usable
gain per stage is to be obtained. Commonly
available tubes which have been designed for
or are suitable for use in video amplifiers are:
6AU6, 6AG5, 6AK5, 6CB6, 6AC7, 6AG7, and
6K6 -GT. Since, at the upper frequency limits
of a video amplifier the input and output
shunting capacitances of the amplifier tubes
have rather low values of reactance, low
values of coupling resistance along with
peaking coils or other special interstage coupling impedances are usually used to flatten
out the gain /frequency and hence the phase/
frequency characteristic of the amplifier.
Recommended operating conditions along with
expressions for calculation of gain and circuit
values are given in figure 9. Only a simple
two -terminal interstage coupling network is
shown in this figure.
The performance and gain -per -stage of a
video amplifier can be improved by the use
of increasingly complex two-terminal inter stage coupling networks or through the use
of four -terminal coupling networks or filters
between successive stages. The reader is referred to Terman's "Radio Engineer's Handbook" for design data on such interstage
coupling networks.
a
Push -Pull Transformer
transformer
coupling between two
stages is illustrated in
figure 10C. This interstage coupling arrangement is fairly commonly used. The system is
particularly effective when it is desired, as in
the system just described, to obtain a fairly
high voltage to excite the grids of a high power audio stage. The arrangement is also
very good when it is desired to apply feedback to the grids of the push -pull stage by
applying the feedback voltage to the lowpotential sides of the two push -pull secondaries.
Impedance coupling between two
stages is shown in figure 10D.
This circuit arrangement is seldom
used, but it offers one strong advantage over
R -C interstage coupling. This advantage is
the fact that, since the operating voltage on
the tube with the impedance in the plate circuit is the plate supply voltage, it is possible
to obtain approximately twice the peak voltage output that it is possible to obtain with
R -C coupling. This is because, as has been
Impedance
Other Interstage
Coupling Methods
Figure 10 illustrates, in addition to resistance- capacitance interstage coupling, seven
additional methods in which coupling between
two successive stages of an audio -frequency
amplifier
may
be
accomplished.
Although
Push -pull
Interstage Coupling
Coupling
6 -7
113
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114
Vacuum
Tube
Amplifiers
THE
RADIO
+e
pA RESISTANCE- CAPACITANCE COUPLING
©
TRANSFORMER COUPLING
©
PUSH -PULL TRANSFORMER COUPLING
©
IMPEDANCE COUPLING
IMPEDANCE -TRANSFORMER COUPLING
0
CATHODE COUPLING
pH
+5
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RESISTANCE- TRANSFORMER COUPLING
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INTERSTAGE
DIR- ECT
Figure 10
COUPLING METHODS FOR AUDIO FREQUENCY VOLTAGE
mentioned before, the d -c plate voltage on an
R -C stage is approximately one -half the plate
supply voltage.
These two circuit arrangements, illustrated
former Coupling
in figures 10E and 10F,
are employed when it is
desired to use transformer coupling for the
reasons cited above, but where it is desired
that the d -c plate current of the amplifier
Impedance -Transformer
and Resistance -Trans-
+5
COUPLING
AMPLIFIERS
stage be isolated from the primary of the coupling transformer. With most types of high permeability wide -response transformers it is
necessary that there be no direct -current flow
through the windings of the transformer. The
impedance- transformer arrangement of figure
10E will give a higher voltage output from
the stage but is not often used since the plate
coupling impedance (choke) must have very
high inductance and very low distributed capacitance in order not to restrict the range of
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HANDBOOK
Phase
Inverters
115
same type tubes with the values of plate voltage and load resistance to be used for the
Gw
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CATHODE RESISTOR
GM OF EACH TUBE
Al
OF EACH TUBE
RP OF EACH TUBE
EQUIVALENT FACTORS INDICATED ABOVE BY (I) ARE
THOSE OBTAINED BY USING AN AMPLIFIER WITH A PAIR
OF SIMILAR TUBE TYPES IN CIRCUIT SHOWN ABOVE.
Figure 11
Equivalent factors for a pair of similar triodes operating as a cathode-coupled audio frequency voltage amplifier.
the transformer which
it
and
its associated
tube feed. The resistance -transformer arrange-
ordinarily quite satisfactory where it is desired to feed a transformer from a voltage amplifier stage with no
d.c.in the transformer primary.
ment of figure 10F is
The cathode coupling arrangement
of figure 10G has been widely used
only comparatively recently. One
outstanding characteristic of such a circuit is
that there is no phase reversal between the
grid and the plate circuit. All other common
types of interstage coupling are accompanied
by a 180° phase reversal between the grid
circuit and the plate circuit of the tube.
Figure 11 gives the expressions for determining the appropriate factors for an equivalent triode obtained through the use of a pair
of similar triodes connected in the cathode coupled circuit shown. With these equivalent
triode factors it is possible to use the expressions shown in figure 5 to determine the
gain of the stage at different frequencies. The
input capacitance of such a stage is less than
that of one of the triodes, the effective grid to -plate capacitance is very much less (it is
so much less that such a stage may be used
as an r -f amplifier without neutralization), and
the output capacitance is approximately equal
to the grid -to -plate capacitance of one of the
triode sections. This circuit is particularly
effective with tubes such as the 6J6, 6N7, and
6SN7 -GT which have two similar triodes in
one envelope. An appropriate value of cathode
resistor to use for such a stage is the value
which would be used for the cathode resistor
cf a conventional amplifier using one of the
Cathode
Coupling
cathode -coupled stage.
Inspection of the equations in figure 11
shows that as the cathode resistor is made
smaller, to approach zero, the Gm approaches
zero, the plate resistance approaches the Rp
of one tube, and the mu approaches zero. As
the cathode resistor is made very large the Gm
approaches one half that of a single tube of
the same type, the plate resistance approaches
twice that of one tube, and the mu approaches
the same value as one tube. But since the Gm
of each tube decreases as the cathode resistor
is made larger (since the plate current will
decrease on each tube) the optimum value of
cathode resistor will be found to be in the
vicinity of the value mentioned in the previous
paragraph.
Direct coupling between successive amplifier stages (plate
of first stage connected directly to the grid of
the succeeding stage) is complicated by the
fact that the grid of an amplifier stage must
be operated at an average negative potential with respect to the cathode of that stage.
However, if the cathode of the second amplifier stage can be operated at a potential more
positive than the plate of the preceding stage
by the amount of the grid bias on the second
amplifier stage, this direct connection between
the plate of one stage and the grid of the succeeding stage can be used. Figure 10H illustrates an application of this principle in
the coupling of a pentode amplifier stage to
the grid of a "hot- cathode" phase inverter. In
this arrangement the values of cathode, screen,
and plate resistor in the pentode stage are
chosen such that the plate of the pentode is at
approximately 0. 3 times the plate supply potential. The succeeding phase- inverter stage
then operates with conventional values of
cathode and plate resistor (same value of resistance) in its normal manner. This type of
phase inverter is described in more detail in
the section to follow.
Direct Coupling
6 -8
Phase Inverters
It is necessary in order to excite the grids
of a push -pull stage that voltages equal in
amplitude and opposite in polarity be applied
to the two grids. These voltages may be obtained through the use of a push -pull input
transformer such as is shown in figure 10C.
It is possible also, without the attendant bulk
and expense of a push -pull input transformer,
to obtain voltages of the proper polarity and
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116
Vacuum Tube
Amplifiers
phase through the use of a so- called phase inverter stage. There are a large number of
phase inversion circuits which have been developed and applied Elut the three shown in
figure 12 have been found over a period of
time to be the most satisfactory from the point
of view of the number of components required
and from the standpoint of the accuracy with
which the two out -of -phase voltages are held
to the same amplitude with changes in supply
voltage and changes in tubes.
All of these vacuum tube phase inverters
are based upon the fact that a 180° phase
shift occurs within a vacuum tube between the
grid input voltage and the plate output voltage.
In certain circuits, the fact that the grid input
voltage and the voltage appearing across the
cathode bias resistor are in phase is used for
phase inversion purposes.
"Hot- Cathode"
Figure 12A illustrates the hot Phase Inverter
cathode type of phase inverter. This type of phase inverter is the simplest of the three types since
it requires only one tube and a minimum of
circuit components. It is particularly simple
when directly coupled from the plate of a
pentode amplifier stage as shown in figure
10H. The circuit does, however, possess the
following two disadvantages: (1) the cathode
of the tube must run at a potential of approximately 0.3 times the plate supply voltage
above the heater when a grounded common
heater winding is used for this tube as well
as the other heater -cathode tubes in a receiver
or amplifier: (2) the circuit actually has a
loss in voltage from its input to either of the
output grids-about 0.9 times the input voltage will be applied to each of these grids.
This does represent a voltage gain of about
1.8 in total voltage output with respect to input (grid -to -grid output voltage) but it is still
small with respect to the other two phase
inverter circuits shown.
Recommended component values for use
with a 6J5 tube in this circuit are shown in
figure 12A. If it is desired to use another tube
in this circuit, appropriate values for the operation of that tube as a conventional amplifier
can be obtained from manufacturer's tube data.
The value of RL obtained should be divided by
two, and this new value of resistance placed
in the circuit as RL. The value of Rk from
tube manual tables should then be used as
Rkl in this circuit, and then the total of Rkl
and Rk2 should be equal to RL.
"Floating Paraphase"
alternate type of
phase inverter sometimes called the "floating paraphase" is illustrated in figure 12B.
This circuit is quite often used with a 6N7
Phase Inverter
An
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Figure 12
THREE POPULAR PHASE -INVERTER CIRCUITS WITH RECOMMENDED VALUES FOR
CIRCUIT COMPONENTS
tube, and appropriate values for the 6N7 tube
in this application are shown. The circuit
shown with the values given will give a voltage gain of approximately 21 from the input
grid to each of the grids of the succeeding
stage. It is capable of approximately 70 volts
peak output to each grid.
The circuit inherently has a small unbalance
in output voltage. This unbalance can be eliminated, if it is required for some special application, by making the resistor Rgl a few per
cent lower in resistance value than RB3.
The circuit shown in figure
12C gives approximately one half the voltage gain from the
input grid to either of the grids of the succeeding stage that would be obtained from a
single tube of the same type operating as a
conventional R -C amplifier stage. Thus, with
a 6SN7 -GT tube as shown (two 6J5's in one
Cathode -Coupled
Phase Inverter
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Vacuum
HANDBOOK
.01
R5
Tube Voltmeter
117
R6
I.R
RP1
D.C.
INPUT
o
Ec
EP
_i11Figure 14
DIRECT COUPLED
D -C
Figure 13
VOLTAGE DIVIDER PHASE
INVERTER
AMPLIFIER
same amplitude as the output of
V
but of
opposite phase.
envelope) the voltage gain from the input
grid to either of the output grids will be approximately 7-the gain is, of course, 14 from
the input to both output grids. The phase
characteristics are such that the circuit is
commonly used in deriving push -pull deflection voltage for a cathode -ray tube from a
signal ended input signal.
The first half of the 6SN7 is used as an
amplifier to increase the amplitude of the applied signal to the desired level. The second
half of the 6SN7 is used as an inverter and
amplifier to produce a signal of the same
amplitude but of opposite polarity. Since the
common cathode resistor, Rk, is not by- passed
the voltage across it is the algebraic sum of
the two plate currents and has the same shape
and polarity as the voltage applied to the input grid of the first half of the 6SN7. When a
signal, e, is applied to the input circuit, the
effective grid- cathode voltage of the first
section is Ae/2, when A is the gain of the
first section. Since the grid of the second
section of the 6SN7 is grounded, the effect of
the signal voltage across Rk (equal to e/2 if
Rk is the proper value) is the same as though
a signal of the same amplitude but of opposite
polarity were applied to the grid. The output
of the second section is equal to Ae /2 if the
plate load resistors are the same for both tube
sections.
commonly used phase inverter is shown in figure 13.
The input section (V,) is connected as a conventional amplifier. The output voltage from V, is impressed on the voltage divider R, -R,. The values of R, and R,
are in such a ratio that the voltage impressed
upon the grid of V2 is 1/A times the output
voltage of V where A is the amplification
factor of V,. The output of Vt is then of the
Voltage Divider
Phase Inverter
A
D -C
6 -9
Amplifiers
Direct current amplifiers are special types
used where amplification of very slow variations in voltage, or of d -c voltages is desired.
amplifier consists of a single
A simple d -c
tube with a grid resistor across the input
terminals, and the load in the plate circuit.
A simple d -c amplifier
circuit is shown in
figure 14, wherein the
the grid of one tube is connected directly to
the plate of the preceding tube in such a
manner that voltage changes on the grid of
the first tube will be amplified by the system.
The voltage drop across the plate coupling
resistor is impressed directly upon the grid
of the second tube, which is provided with
enough negative grid bias to balance out the
excessive voltage drop across the coupling
resistor. The grid of the second tube is thus
maintained in a slightly negative position.
The d -c amplifier will provide good low frequency response, with negligible phase distortion. high frequency response is limited
by the shunting effect of the tube capacitances,
as in the normal resistance coupled amplifier.
A common fault with d -c amplifiers of all
types is static instability. Small changes in
the filament, plate, or grid voltages cannot
be distinguished from the exciting voltage.
Regulated power supplies and special balancing circuits have been devised to reduce the
effects of supply variations on these amplifiers. A successful system is to apply the
plate potential in phase to two tubes, and to
apply the exciting signal to a push -pull grid
Basic
D -C
Amplifier Circuit
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118
Vacuum
Amplifiers
Tube
THE
RADIO
BALANCE
CONTROL
Figure
15
LOFTIN -WHITE
D -C AMPLIFIER
Figure
16
PUSH -PULL D -C AMPLIFIER
WITH EITHER SINGLE -ENDED
OR PUSH -PULL INPUT
circuit configuration. If the two tubes are
identical, any change in electrode voltage is
balanced out. The use of negative feedback
can also greatly reduce drift problems.
The
"Loftin -Whiter
Circuit
Two
stages
amplifier
d -c
may be arranged,
so that their plate
supplies are effectively in series, as illustrated in figure 15. This is known as a Loftin White amplifier. All plate and grid voltages
may be obtained from one master power supply
instead of separate grid and plate supplies.
A push-pull version of this amplifier (figure 16)
can be used to balance out the effects of slow
variations in the supply voltage.
6 -10
Single -ended Triode
Amplifiers
Figure 17 illustrates five circuits for the
operation of Class A triode amplifier stages.
Since the cathode current of a triode Class Al
(no grid current) amplifier stage is constant
with and without excitation, it is common
practice to operate the tube with cathode
bias. Recommended operating conditions in
regard to plate voltage, grid bias, and load
impedance for conventional triode amplifier
stages are given in the RCA Tube Manual,
RC -16.
It is possible, under certain
conditions to operate singleended triode amplifier stages
pentode and tetrode stages as well) with
excitation of sufficient amplitude that
current is taken by the tube on peaks.
type of operation is called Class A2 and
Extended Class A
Operation
(and
grid
grid
This
is characterized by increased plate -circuit
efficiency over straight Class A amplification
without grid current. The normal Class A1
amplifier power stage will operate with a plate
circuit efficiency of from 20 per cent to perhaps
35 per cent. Through the use of Class A2
operation it is possible to increase this plate
circuit efficiency to approximately 38 to 45
per cent. However, such operation requires
careful choice of the value of plate load impedance, a grid bias supply with good regulation (since the tube draws grid current on
peaks although the plate current does not
change with signal), and a driver tube with
moderate power capability to excite the grid
of the Class A2 tube.
Figures 17D and 17E illustrate two methods
of connection for such stages. Tubes such as
the 845, 849, and 304TL are suitable for such
a stage. In each case the grid bias is approximately the same as would be used for a Class
Al amplifier using the same tube, and as
mentioned before, fixed bias must be used
-
along with an audio driver of good regulation
preferably a triode stage with a 1:1 or step down driver transformer. In each case it will
be found that the correct value of plate load
impedance will be increased about 40 per cent
over the value recommended by the tube manufacturer for Class A1 operation of the tube.
Class A power amplifier
operates in such a way as
to amplify as faithfully as
possible the waveform applied to the grid of the tube. Large power output is of more importance than high voltage
amplification, consequently gain characteristics may be sacrificed in power tube design
to obtain more important power handling capabilities. Class A power tubes, such as the 45,
2A3 and 6ÁS7 are characterized by a low
amplification factor, high plate dissipation
and relatively high filament emission.
The operating characteristics of a Class A
Operation Characteristics of a Triode
Power Amplifier
www.americanradiohistory.com
A
Triode Amplifier Characteristics
HANDBOOK
E5
-(0.
119
68 x Ebb)
ll
There Ebb is the actual plate voltage of
the Class A stage, and µ is the amplification factor of the tube.
pA IMPEDANCE
3-
COUPLING
4-
5-
®
TRANSFORMER COUPLING
6-
7-
©
IMPEDANCE -TRANSFORMER COUPLING
8-
Locate the E5 bias point on the IP vs.
Ep graph where the E5 bias line crosses
the plate voltage line, as shown in figure
18. Call this point P.
Locate on the plate family of curves the
value of zero -signal plate current, I
corresponding to the operating point, P.
Locate 2 x 1, (twice the value of 1p) on
the plate current axis (Y- axis). This
point corresponds to the value of maximum signal plate current, imu.
Locate point x on the d -c bias curve at
zero volts (Eg = 0), corresponding to the
value of imax.
Draw a straight line(x - y) through points
x and P. This line is the load resistance
line. Its slope corresponds to the value
of the load resistance.
Load Resistance, (in ohms)
RL
Zsz RL
-
emu
.
¡max
- Cain
- tmin
where e is in volts, i is in amperes, and
RL is in ohms.
-e1AS
=
0 TRANSFORMER COUPLING
At OPERATION
-
AUTO
TRANSFORMER
-DIAS
©
e+e
TO
CLASS C
LOAD
CLASS Az MODULATOR WITH AUTO-TRANSFORMER COUPLING
Figure
17
Output coupling arrangements for single -ended
Class A triode audio -frequency power amplifiers.
triode amplifier employing an output transformer- coupled load may be calculated from
the plate family of curves for the particular
tube in question by employing the following
steps:
1- The load resistance should be approximately twice the plate resistance of the
tube for maximum undistorted power output. Remember this fact for a quick check
calculations.
Calculate the zero -signal bias voltage
on
2-
(Eg).
Multiply the zero - signal plate
current, Ii,, by the operating plate voltage, Ep. If the plate dissipation rating
of the tube is exceeded, it is necessary
to increase the bias (E5) on the tube so
that the plate dissipation falls within
the maximum rating of the tube. If this
step is taken, operations 2 through 8
must be repeated with the new value of
9- Check:
+e
FOR
E5.
10- For maximum power output, the peak a -c
grid voltage on the tube should swing to
2E5 on the negative cycle, and to zero bias on the positive cycle. At the peak
of the negative swing, the plate voltage reaches emu and the plate current
drops to iain On the positive swing of
the grid signal, the plate voltage drops
to envia and the plate current reaches
ima. The power output of the tube is:
Power Output (watts)
Po -
(imax
-
¡min) x (emaz
- emin)
8
where i is in amperes and
e
is in volts.
11- The second harmonic distortion generated
in a single -ended Class A triode amplifier, expressed as a percentage of the
fundamental output signal is:
www.americanradiohistory.com
120
250
MN111
to
W200 SOW.
cc
7
.
W
ai11
,50
a
RADIO
ou
f
:
THE
Amplifiers
Tube
11
ä
_
a1
11
I71
gIf/
I\
Vacuum
Figure
19
Normal single -ended pentode or beam tetrad.
audio- frequency power output stage.
xi
I.1
IMIN -
0
v
-
I
200
EMIN.
300
EG
PLATE VOLTS
400
1
EMAX.
AVERAGE PLATE CHARACTERISTICS
p. =4.2
Rp
/
/
1
100
- 2A3
OHMS
PLATE DISSIPATION =15 WATTS
=
BOO
LOAD RESISTANCE
RL
-
EMAZ
I
-
EMIN.
MAX.' IMIN.
OHMS
POWER OUTPUT
Po
(IMAX.
-IMIN) IEMAX- Ei,) WATTS
8
- IP
X
'MAX.
Figure
100 PERCENT
18
Formulas for determining the operating conditions for a Class A triode single -ended audio frequency power output stage. A typical load
line has been drawn on the average plate characteristics of a type 2A3 tube to illustrate
the procedure.
%
0. 9 Ep
2d harmonic =
(imax
- imin)
IP
iman
and the power output is somewhat less than
Ip
2
(x
Ep x Ip
100)
-imin
2
Figure 18 illustrates the above steps as
applied to a single Class A 2A3 amplifier stage.
6 -11
The operating characteristics of pentode power
amplifiers may be obtained
from the plate family of
curves, much as in the manner applied to
triode tubes. A typical family of pentode plate
curves is shown in figure 20. It can be seen
from these curves that the plate current of the
tube is relatively independent of the applied
plate voltage, but is sensitive to screen voltage. In general, the correct pentode load resistance is about
Operating Characteristics of a Pentode
Power Amplifier
SECOND HARMONIC DISTORTION
IMAX.+ IMiN.)
2
plifier stage. Tubes of this type have largely
replaced triodes in the output stage of receivers and amplifiers due to the higher plate
efficiency (30 % -40 %) with which they operate.
Tetrode and pentode tubes do, however, introduce a considerably greater amount of harmonic
distortion in their output circuit, particularly
odd harmonics. In addition, their plate circuit
impedance (which acts in an amplifier to damp
loudspeaker overshoot and ringing, and acts
in a driver stage to provide good regulation) is
many times higher than that of an equivalent
triode. The application of negative feedback
acts both to reduce distortion and to reduce
the effective plate circuit impedance of these
tubes.
Single -ended Pentode
Amplifiers
Figure 19 illustrates the conventional circuit for a single -ended tetrode or pentode am-
These formulae may be used for a quick check
on more precise calculations. To obtain the
operating parameters for Class A pentode amplifiers, the following steps are taken:
1- The imax point is chosen so as to fall on
the zero -bias curve, just above the
"knee" of the curve (point A, figure 20) .
2- A preliminary operating point, P, is determined by the intersection of the plate
voltage line, Fp, and the line of imaz /2.
www.americanradiohistory.com
Push -Pull
'HANDBOOK
Amplifiers
121
is:
Power Output (watts)
6- The power output
(¡max
.
,iAA
Po0
.
.OAD_iNE
t4/. ,OA^I¡vE
AP=PB
+ 1.41
eR
%
P(STATIC VALUE)
eMA%
is equal to:
¡max
¡max
Where IP
Figure 20
%
POWER AMPLIFIER
is the negative control grid voltage at
the operating point P
The grid voltage curve that this point
falls upon should be one that is about a
the value of ER required to cut the plate
current to a very low value (Point B).
Point B represents imin on the plate current axis (y-axis). The line ima /2 should
be located half-way between ima and
twin
trial load line is constructed about
point P and point A in such a way that
the lengths A -P and P -B are approximately equal.
hen the most satisfactory load line has
4been determined, the load resistance may
calculated:
3- A
emax
-
envia
imax
-
imin
X
RL
ER + 0. 7 F.R.
distortion is:
distortion
-
min
-
1,
2
x100
(Ix - Iy)
- ¡min
is the static plate current of
+
1.41
of the tube.
GRAPHIC DETERMINATION OF OPERATING CHARACTERISTICS OF A PENTODE
RL
2d harmonic
=
PLATE VOLTS
"V"
-4)2
Where
is the plate current at the point
on the load line where the grid voltage,
eR, is equal to: ER - 0. 7 F.R; and where
Iy is the plate current at the point where
7- The percentage harmonic
e MIN
(Ix
32
l
CHOOSE
SOrHAT
- ¡min)
5- The operating bias (ER) is the bias at
point P.
3d harmonic
(Ix
- Ty)
+ 1.41 (1x
- ly)
¡max
-imin -1.41
¡max
- tmin
6 -12
x100
Push -Pull Audio
Amplifiers
A number of advantages are obtained through
the use of the push -pull connection of two or
four tubes in an audio -frequency power amplifier. Two conventional circuits for the use
of triode and tetrode tubes in the push -pull
connection are shown in figure 21. The two
main advantages of the push -pull circuit arrangement are: (1) the magnetizing effect
of the plate currents of the output tubes is
cancelled in the windings of the output transformer; (2) even harmonics of the input signal
(second and fourth harmonics primarily) generated in the push -pull stage are cancelled
when the tubes are balanced.
The cancellation of even harmonics generated in the stage allows the tubes to be oper-
PUSH -PULL TRIODE AND TETRODE
FIGURE
distortion
21
www.americanradiohistory.com
122
Ii
e..
itdNii,
I.a
r11\I,11
Vacuum
300
Amplifiers
Tube
THE
J111111rrI
RADIO
1111NN1111NNN11NNlli' NN111111.
Ilun
.rv
IIM
i
iiii:iii:'
:
11N
/
I
.
.
1/
"um=
iGGiiGiiGiiG
rM\RJ111N.
1111N1C::NII::..Ci
'
I%
11111111NII Ci7 Ci
GGG
táiGGiiGiï
N..llrCiC
1111I
r
A
250
,
11I.1111AIi
11
,
VALUE Of
ZERO SIGNAL
PLATE CUR
N/
W/N
M
Y w1'CCl
M. :íRG
/N!!:í.
t1111P..
50
111/111/11
!!I!!1!!1!!IIlII!I!!a 111111111111111
111111111111111111,¡m1111111111N111N
11111111111111111i1111111111111111111111
1111111111111111i/1111111111.1111111111
Iiill iiiiiii%1/N1111111111111111111
1111111111111
IIIIIIN11/I
1111111111
IIIINIIII111111
i'Il:il:?'i11NN11111111111111111111111
..\!
PLATE VOLTS
11
11111
1
1111111111
Cil:i=l. 'MIN_
00
.1!
-.
II74VU'A
l....
11
o.
I.M
II
11111 P;11161111N111111111111111111111
1l:í1111D41N1 111111111111111111111111
70
e0 -30 -<0
30 -20
-0
0
-60
300
(EP)
GRID VOLTS (EG)
Figure 22
DETERMINATION OF OPERATING PARAMETERS
FOR PUSH-PULL CLASS A
TRIODE TUBES
-in
ated Class AB
other words the tubes may
be operated with bias and input signals of
such amplitude that the plate current of alternate tubes may be cut off during a portion of
the input voltage cycle. If a tube were operated
in such a manner in a single -ended amplifier
the second harmonic amplitude generated would
be prohibitively high.
Push -pull Class AB operation allows a plate
circuit efficiency of from 45 to 60 per cent to
be obtained in an amplifier stage depending
upon whether or not the exciting voltage is
of such amplitude that grid current is drawn
by the tubes. If grid current is taken on input
voltage peaks the amplifier is said to be operating Class AB2 and the plate circuit efficiency can be as high as the upper value just
mentioned. If grid current is not taken by the
stage it is said to be operating Class AB1 and
the plate circuit efficiency will be toward the
lower end of the range just quoted. In all Class
AB amplifiers the plate current will increase
from 40 to 150 per cent over the no- signal
value when full signal is applied.
The operating characteristics of push pull Class A amplifiers may also be
determined from the plate family of curves for
Operating Characteristics
of Push -Pull Class A
Triode Power Amplifier
a
particular triode tube by the following steps:
1- Erect a vertical line from the plate voltage axis (x -axis) at 0.6 Ep (figure 22),
which intersects the Eg = 0 curve. This
point of intersection (P), interpolated to
the plate current axis (y -axis) may be
taken as imp. It is assumed for simplification that imaz occurs at the point of
the zero -bias curve corresponding to
2-
0.6 Ep.
The power output obtainable from the two
tubes is:
Power output (Po)
-
i
x Ep
5
where PO is expressed in watts, imax in
amperes, and Ep is the applied plate
voltage.
3-
a preliminary load line through
point P to the Ep point located on the
x -axis (the zero plate current line). This
load line represents % of the actual plate to -plate load of the Class A tubes. Therefore:
Draw
RL
(plate -to- plate)
www.americanradiohistory.com
Ep
= 4 x
1.6 ED
max
0.6 Ep
¡Max
Class
I-ANDBOOK
where RL is expressed in ohms, Ep in
volts, and imu in amperes.
Figure 22 illustrates the above steps applied to a push -pull Class A amplifier using
two 2A3 tubes.
4- The average plate current is 0.636 Imp,
and, multiplied by the plate voltage, Ep,
will give the average watts input to the
plates of the two tubes. The power output should be subtracted from this value
to obtain the total operating plate dissipation of the two tubes. If the plate
dissipation
is excessive, a slightly
higher value of RL should be chosen to
limit the plate dissipation.
5- The correct value of operating bias, and
the static plate current for the push -pull
tubes may be determined from the Eg vs.
1p curves, which are a derivation of the
Ep vs. Ip curves for various values of
Eg.
6- The Fg vs. Ip curve may be constructed
in this manner: Values of grid bias are
read from the intersection of each grid
bias curve with the load line. These
points are transferred to the Eg vs. Ip
graph to produce a curved line, A -B. If
the grid bias curves of the Ep vs. Ip
graph were straight lines, the lines of
the Eg vs. 1p graph would also be straight
This is usually not the case. A tangent
to this curve is therefore drawn, starting
at point A', and intersecting the grid
voltage abscissa (x- axis). This intersection (C) is the operating bias point
for fixed bias operation.
7- This operating bias point may now be
plotted on the original Eg vs. 1p family
of curves (C'), and the zero-signal current produced by this bias is determined.
This operating bias point (C') does not fall
on the operating load line, as in the case of a
single -ended amplifier.
8- Under conditions of maximum power output, the exciting signal voltage swings
from zero-bias voltage to zero -bias voltage for each of the tubes on each half
of the signal cycle. Second harmonic
distortion is largely cancelled out.
6
-13
B Audio Frequency
Power Amplifiers
Class
The Class B audio- frequency power amplifier (figure 23) operates at a higher plate circuit efficiency than any of the previously
described types of audio power amplifiers.
Full- signal plate- circuit efficiencies of 60 to
B
Bt
Audio
DRIVER
Amplifiers
- BIAS
Bt
123
MOD
(GROUND FOR
ZERO WAS
OPERATING
CONDITION)
Figure 23
CLASS B AUDIO FREQUENCY
POWER AMPLIFIER
70 per cent are readily obtainable with the
tube types at present available for this type
of work. Since the plate circuit efficiency is
higher, smaller tubes of lower plate dissipation may be used in a Class B power amplifier
of a given power output than can be used in
any other conventional type of audio amplifier.
An additional factor in favor of the Class B
audio amplifier is the fact that the power input to the stage is relatively low under nosignal conditions. It is for these reasons that
this type of amplifier has largely superseded
other types in the generation of audio -frequency
levels from perhaps 100 watts on up to levels
of approximately 150,000 watts as required for
large short -wave broadcast stations.
Disadvantages of
Class B Amplifier
There are attendant dis advantageous features to the
Operation
operation of a power amplifier of this type; but all
these disadvantages can be overcome by proper
design of the circuits associated with the
power amplifier stage. These disadvantages
are: (1) The Class B audio amplifier requires
driving power in its grid circuit; this disadvantage can be overcome by the use of an
oversize power stage preceding the Class B
stage with a step -down transformer between
the driver stage and the Class -B grids. Degenerative feedback is sometimes employed
to reduce the plate impedance of the driver
stage and thus to improve the voltage regulation under the varying load presented by the
Class B grids. (2) The Class B stage requires
a constant value of average grid bias to be
supplied in spite of the fact that the grid current on the stage is zero over most of the
cycle but rises to values as high as one -third
of the peak plate current on the peak of the
exciting voltage cycle. Special regulated bias
supplies have been used for this application,
or B batteries can be used. However, a number
www.americanradiohistory.com
124
Vacuum Tube
Amplifiers
THE
of tubes especially designed for Class B audio
amplifiers have been developed which require
zero average grid bias for their operation. The
811A, 838, 805, 809, HY -5514, and TZ -40 are
examples of this type of tube. All these so-
called "zero- bias" tubes have rated operating
conditions up to moderate plate voltages
wherein they can be operated without grid
bias. As the plate voltage is increased to
to their maximum ratings, however, a small
amount of grid bias, such as could be obtained
from several 4 1/2-volt C batteries, is required.
(3), A Class B audio -frequency power amplifier or modulator requires a source of plate
supply voltage having reasonably good regulation. This requirement led to the development
of the swinging choke. The swinging choke is
essentially a conventional filter choke in
which the core air gap has been reduced. This
reduction in the air gap allows the choke to
have a much greater value of inductance with
low current values such as are encountered
with no signal or small signal being applied
to the Class B stage. With a higher value of
current such as would be taken by a Class B
stage with full signal applied the inductance
of the choke drops to a much lower value.
With a swinging choke of this type, having
adequate current rating, as the input inductor
in the filter system for a rectifier power supply, the regulation will be improved to a point
which is normally adequate for a power supply
for a Class B amplifier or modulator stage.
Calculation of Operating
Conditions of Class B
Power Amplifiers
The following procedure can be used for
the calculation of the
operating conditions
of Class B power amplifiers when they are to
operate into a resistive load such as the type
of load presented by a Class C power amplifier. This procedure will be found quite satisfactory for the application of vacuum tubes as
Class B modulators when it is desired to
operate the tubes under conditions which are
not specified in the tube operating characteristics published by the tube manufacturer. The
same procedure can be used with equal effectiveness for the calculation of the operating
conditions of beam tetrodes as Class AB2
amplifiers or modulators when the resting
plate current on the tubes (no signal condition) is less than 25 or 30 per cent of the
maximum -signal plate current.
1- With the average plate characteristics
of the tube as published by the manufacturer before you, select a point on
the Ep = E& (diode bend) line at about
twice the plate current you expect the
tubes to kick to under modulation. If
beam tetrode tubes are concerned, select
RADIO
a point at about the same amount of plate
current mentioned above, just to the
right of the region where the Ib line
takes a sharp curve downward. This will
be the first trial point, and the plate
voltage at the point chosen should be
not more than about 20 per cent of the
d -c voltage applied to the tubes if good
plate- circuit efficiency is desired.
Note down the value of ipp. and cp.,¡, at
this point.
3- Subtract the value of epm¡ from the d -c
plate voltage on the tubes.
4- Substitute the values obtained in the
following equations:
2-
P0
=
pmau(Ebb
RL_4 (Ebb
epmin)
=
Power output
from 2 tubes
emu.)
i pma:
=
Plate -to -plate load
for
2
tubes
Full signal efficiency (Nu)
78.5
Cl_evm
Ebb
/I
Effects of Speech All the above equations are
Clipping
true for sine -wave operating
conditions of the tubes concerned. However, if a speech clipper is being
used in the speech amplifier, or if it is desired
to calculate the operating conditions on the
basis of the fact that the ratio of peak power
to average power in a speech wave is approximately 4 -to-1 as contrasted to the ratio of
2 -to-1 in a sine wave-in other words, when
non- sinusoidal waves such as plain speech or
speech that has passed through a clipper are
concerned, we are no longer concerned with
average power output of the modulator as far
as its capability of modulating a Class -C amplifier is concerned; we are concerned with its
peak -power- output capability.
Under these conditions we call upon other,
more general relationships. The first of these
is: It requires a peak power output equal to
the Class -C stage input to modulate that input
fully.
The second one is: The average power output required of the modulator is equal to the
shape factor of the modulating wave multiplied by the input to the Class -C stage. The
shape factor of unclipped speech is approximately 0. 25. The shape factor of a sine wave
is 0. 5. The shape factor of a speech wave that
www.americanradiohistory.com
eoo
ï
ó
600
-111
ma.
U1
U
Ò
400
125
200
O.C.
-
o
N(Ong_.
H-201111.
agarla...
- +6a
vaLTs Ecc
p10
N'
been drawn
on the overage
characteristics of o type 811
tube.
Parameters
B
EF e 6.3 VOLTS
.u.
Figure 24
Typical Class 8 o-f amplifier
load line. The load line has
m
d'
n
NA
:Ma
,C
W! ..s/tll
Class
HANDBOOK
11iáse
-O=e=sr =
s
I
400
600
1200
_
2400
2000
1800
PLATE VOLTS (Ebb)
AVERAGE PLATE CHARACTERISTICS TYPE 811 AND 811 -A
has been passed through a clipper -filter arrangement is somewhere between 0. 25 and 0. 9
depending upon the amount of clipping that
has taken place. With 15 or 20 db of clipping
the shape factor may be as high as the figure
of 0.9 mentioned above. This means that the
audio power output of the modulator will be
90% of the input to the Class -C stage. Thus
with a kilowatt input we would be putting
900 watts of audio into the Class -C stage for
100 per cent modulation as contrasted to perhaps 250 watts for unclipped speech modulation of 100 per cettt.
Figure 24 shows a set of
plate characteristics for
a type 811A tube with a
load line for Class B operation. Figure 25
lists a sample calculation for determining the
proper operating conditions for obtaining approximately 185 watts output from a pair of
the tubes with 1000 volts d -c plate potential.
Also shown in figure 25 is the method of determining the proper ratio for the modulation
transformer to couple between the 811's or
811A's and the anticipated final amplifier
which is to operate at 2000 plate volts and
175 ma. plate current.
Sample Calculation
for 811A Tubes
Lion shown in figure 25. or by reference to the
published characteristics on the tubes to be
used. (2) Determine the load impedance which
will be presented by the Class C amplifier
stage to be modulated by dividing the operating
plate voltage on that stage by the operating
value of plate current in amperes. (3) Divide
the Class C load impedance determined in (2)
SAMPLE CALCULATION
CONDITION:
2 TYPE 811 TUBES, Ebb, = 1000
INPUT TO FINAL STAGE, 350 W.
PEAR POWER OUTPUT NEEDED. 350 IS% = 370
FINAL AMPLIFIER Ebb = 2000 V.
FINAL AMPLIFIER Ib = .175 A.
FINAL AMPLIFIER ZL = -22SISL = 11400 R
.175
EXAMPLE
CHOSE POINT ON 811
TO RIGHT OF
IP
MAX.
IG MAX.
PEAK PO
Ebb'
F /G.
EP MIN.
A.
EG MAX. _
RL
=
4 X
NP
=
78.5 (1
.410
_
=
X
900
-
)
1
(.9)
76.5
=
WO (AVERAGE WITH SINE WAVE)
WIN
=
- 260
Ió.5
Ib (MAXIMUM
WO PEAR
=
=
369
W.
=
X80
=
=
70.5 "b
POIPEAR)_I813W
W.
WITH SINE WAVE)
100
DRIVING POWER
80
8800 n.
=
:9000
24 )
+100
A.
.100
.410 0 (1000 -10o)
=
CHARACTERISTICS JUST
Ecc. (PO /NT X.
=.410
_
W.
=
=
260 MA
e W
WZ PR
-
W.
TRANSFORMER:
The method illustrated
in figure 25 can be used
in general for the determination of the proper transformer ratio to
couple between the modulator tube and the
amplifier to be modulated. The procedure can
be stated as follows: (1) Determine the proper
plate -to-plate load impedance for the modulator
tubes either by the use of the type of calculaModulation Transformer
Calculation
114
ZP
sew
TURNS RATIO
=
- 1.29
LA
ZP
=
1
29
=
1.14 STEP UP
Figure 25
Typical calculation of operating conditions for
a Class B a -f power amplifier using a pair of
type 811 or 811A tubes. Plate characteristics
and load line shown in figure 24.
www.americanradiohistory.com
126
Vacuum
Tube
Amplifiers
above by the plate -to -plate load impedance for
the modulator tubes determined in (1) above.
The ratio determined in this way is the sec ondary-to- primary impedance ratio. (4) Take
the square root of this ratio to determine the
secondary-to- primary turns ratio. If the turns
ratio is greater than one the use of a step -up
transformer is required. If the turns ratio as
determined in this way is less than one a stepdown transformer is called for.
If the procedure shown in figure '25 has
been used to calculate the operating conditions
for the modulator tubes, the transformer ratio
calculation can be checked in the following
manner: Divide the plate voltage on the modulated amplifier by the total voltage swing on
the modulator tubes: 2 (Ebb
e, 0). This ratio
should be quite close numerically to the transformer turns ratio as previously determined.
The reason for this condition is that the ratio
between the total primary voltage and the d-c
plate supply voltage on the modulated stage
is equal to the turns ratio of the transformer,
since a peak secondary voltage equal to the
plate voltage on the modulated stage is required to modulate this stage 100 per cent.
-
Use of Clipper Speech
Amplifier with Tetrode
Modulator Tubes
clipper speech
amplifier is used in
conjunction with a Class
current.
As stated previously, a
Class B audio amplifier
requires the driving stage
to supply well -regulated audio power to the
grid circuit of the Class B stage. Since the
performance of a Class B modulator may easily
be impaired by an improperly designed driver
stage, it is well to study the problems incurred in the design of the driver stage.
The grid circuit of a Class B modulator may
be compared to a variable resistance which
decreases in value as the exciting grid voltage is increased. This variable resistance appears across the secondary terminals of the
driver transformer so that the driver stage is
Class
B
Modulators
RADIO
called upon to deliver power to a varying load.
For best operation of the Class 13 stage, the
grid excitation voltage should not drop as the
power taken by the grid circuit increases.
These opposing conditions call for a high order of voltage regulation in the driver stage
plate circuit. In order to enhance the voltage
regulation of this circuit, the driver tubes must
have low plate resistance, the driver transformer must have as large a step -down ratio
as possible, and the d-c resistance of both
primary and secondary windings of the driver
transformer should be low.
The driver transformer should reflect into
the plate circuit of the driver tubes a load of
such value that the required driving power is
just developed with full excitation applied to
the driver grid circuit. If this is done, the
driver transformer will have as high a stepdown ratio as is consistent with the maximum
drive requirements of the Class B stage. If
the step -down ratio of the driver transformer is
too large, the driver plate load will be so
high that the power required to drive the Class
B stage to full output cannot be developed.
If the step-down ratio is too small the regulation of the driver stage will be impaired.
When a
B modulator stage, the
plate current on that stage will kick to a
higher value with modulation(due to the greater
average power output and input) but the plate
dissipation on the tubes will ordinarily be
less than with sine -wave modulation. However,
when tetrode tubes are used as modulators,
the screen dissipation will be much greater
than with sine -wave modulation. Care must
be taken to insure that the screen dissipation rating on the modulator tubes is not exceeded under full modulation conditions with
a clipper speech amplifier. The screen dissipation is equal to screen voltage times screen
Practical Aspects of
THE
Driver Stage
Calculations
The parameters for the driver
stage may be calculated from
the plate characteristic curve, a
sample of which is shown in figure 24. The
required positive grid voltage (eg -m,$) for the
811A tubes used in the sample calculation is
found at point X, the intersection of the load
line and the peak plate current as found on the
y -axis.
This is + 80 volts. If a vertical line
is dropped from point X to intersect the dotted
current curves, it will be found that the
current for a single 811A at this value of
voltage is 100 milliamperes (point Y).
peak grid driving power is therefore
80 x 0.100 = 8 watts. The approximate average
driving power is 4 watts. This is an approximate figure because the grid impedance is not
constant over the entire audio cycle.
A pair of 2A3 tubes will be used as drivers,
operating Class A, with the maximum excitation to the drivers occuring just below the
point of grid current flow in the 2A3 tubes.
The driver plate voltage is 300 volts, and the
grid bias is -62 volts. The peak power developed in the primary winding of the driver
transformer is:
grid
grid
grid
The
Peak Power (Pe) = 2R1 I
PE&
,a
`Rp + RIwhere it is the amplification factor of the
driver tubes (4.2 for 2A3). Eg is the peak grid
swing of the driver stage (62 volts). Rp is the
(watts)
www.americanradiohistory.com
Cathode Follower Amplifier
HANDBOOK
plate resistance of one driver tube (800 ohms).
RL is % the plate -to -plate load of the driver
stage, and Pp is 8 watts.
Solving the above equation for RL, we
obtain a value of 14,500 ohms load, plate -toplate for the 2A3 driver tubes.
The peak primary voltage is:
epri = 2RL x
g
Ft,
+RL
493 volts
and the turns ratio of the driver transformer
(primary to % secondary) is:
epri
-=
493
=
eg(ma:)
80
6.15:1
Plate Circuit One of the commonest
distortion in a Class
Impedance
causes of
B
modu-
lator is incorrect load impedance
in the plate circuit. The purpose
of the Class B modulation transformer is to
take the power developed by the modulator
(which has a certain operating impedance) and
transform it to the operating impedance imposed by the modulated amplifier stage.
If the transformer in question has the same
number of turns on the primary winding as it
has on the secondary winding, the turns ratio
is 1:1, and the impedance ratio is also 1:1. If
a 10,000 ohm resistor is placed across the
secondary terminals of the transformer, a reflected load of 10,000 ohms would appear
across the primary terminals. If the resistor
is changed to one of 2376 ohms, the reflected
primary impedance would also be 2376 ohms.
If the transformer has twice as many turns
on the secondary as on the primary, the turns
ratio is 2:1. The impedance ratio is the square
of the turns ratio, or 4:1. If a 10,000 ohm
resistor is now placed across the secondary
winding, a reflected load of 2,500 ohms will
appear across the primary winding.
Matching
It can be seen
from the
above paragraphs that the
Class B modulator plate
load is entirely dependent upon the load
placed upon the secondary terminals of the
Class B modulation transformer. If the secondary load is incorrect, certain changes will
take place in the operation of the Class B
modulator stage.
When the modulator load impedance is too
low, the efficiency of the Class B stage
is reduced and the plate dissipation of the
tubes is increased. Peak plate current of
the modulator stage is increased, and saturation of the modulation transformer core may
result. "Talk -back" of the modulation trans-
Effects of Plate
Circuit Mis -match
127
former may result if the plate load impedance
of the modulator stage is too low.
When the modulator load impedance is too
high, the maximum power capability of the
stage is reduced. An attempt to increase the
output by increasing grid excitation to the
stage will result in peak -clipping of the audio
wave. In addition, high peak voltages may be
built up in the plate circuit that may damage
the modulation transformer.
6 -14
Cathode- Follower
Power Amplifiers
The cathode -follower is essentially a power
output stage in which the exciting signal is
applied between grid and ground. The plate is
maintained at ground potential with respect to
input and output signals, and the output signal
is taken between cathode and ground.
Figure 26 illustrates four
types of cathode - follower
power amplifiers in common usage and figure 27 shows the output
impedance (Ro), and stage gain (A) of both
triode and pentode(or tetrode) cathode- follower
stages. It will be seen by inspection of the
equations that the stage voltage gain is always
less than one, that the output impedance of
the stage is much less than the same stage
operated as a conventional cathode -return
amplifier. The output impedance for conventional tubes will be somewhere between
100 and 1000 ohms, depending primarily on
the transconductance of the tube.
This reduction in gain and output impedance for the cathode -follower comes about
since the stage operates as though it has 100
per cent degenerative feedback applied between
its output and input circuit. Even though the
voltage gain of the stage is reduced to a value
less than one by the action of the degenerative
feedback, the power gain of the stage (if it is
operating Class A) is not reduced. Although
more voltage is required to excite a cathode follower amplifier than appears across the load
circuit, since the cathode "follows" along
with the grid, the relative grid-to- cathode voltage is essentially the same as in a conventional amplifier.
Types of Cathode-
Follower Amplifiers
Although the cathode -follower gives no voltage
gain, it is an effective
power amplifier where it is desired to feed a
low- impedance load, or where it is desired to
feed a load of varying impedance with a signal
having good regulation. This latter capability
Use of Cathode-
Follower Amplifiers
www.americanradiohistory.com
128
Vacuum
Tube
Amplifiers
THE
TRIODE
-U
ucr
J,1
+1
Re (CATHODE
PENTODE:
Ro(cAr.,00E
A
=
RADIO
A
+
GM
RL
L
+Rp
(Rn,+Rea)
RK,
Rao
RL
)
RL(.U+I
Ri
+Rn2+ RL'
R
1+RL Gu
G.. Rea
Figure 27
Equivalent
factors for pentode (or tetrad.)
cathode- follower power amplifiers.
plifier tube, the components
Figure 26
CATHODE-FOLLOWER OUTPUT
CIRCUITS FOR AUDIO OR
VIDEO AMPLIFIERS
makes the cathode follower particularly effective as a driver for the grids of a Class B
modulator stage.
The circuit of figure 26A is the type of amplifier, either single -ended or push -pull, which
may be used as a driver for a Class B modulator or which may be used for other applications such as feeding a loudspeaker where unusually good damping of the speaker is desired. If the d -c resistance of the primary of
the transformer T2 is approximately the correct
value for the cathode bias resistor for the am-
Rk and Ck need
not be used. Figure 26B shows an arrangement
which may be used to feed directly a value of
load impedance which is equal to or higher
than the cathode impedance of the amplifier
tube. The value of Cc must be quite high,
somewhat higher than would be used in a conventional circuit, if the frequency response of
the circuit when operating into a low- impedance load is to be preserved.
Figures 26C and 26D show cathode -follower
circuits for use with tetrode or pentode tubes.
Figure 26C is a circuit similar to that shown
in 26A and essentially the same comments
apply in regard to the components Rk and Ck
and the primary resistance of the transformer
T2. Notice also that the screen of the tube is
maintained at the same signal potential as the
cathode by means of coupling capacitor Cd.
This capacitance should be large enough so
that at the lowest frequency it is desired to
pass through the stage its reactance will be
low with respect to the dynamic screen -tocathode resistance in parallel with Rd T2 in
this stage as well as in the circuit of figure
26A should have the proper turns (or impedance) ratio to give the desired step -down or
step -up from the cathode circuit to the load.
Figure 26D is an arrangement frequently used
in video systems for feeding a coaxial cable of
relatively low impedance from a vacuum -tube
amplifier. A pentode or tetrode tube with a
cathode imped*tce as a cathode follower
(1 /G,a) approximately the same as the cable
impedance should be chosen. The 6AG7 and
6AC7 have cathode impedances of the same
order as the surge impedances of certain types
of low- capacitance coaxial cable. An arrangement such as 26D is also usable for feeding
coaxial cable with audio or r -f energy where
it is desired to transmit the output signal
over moderate distances. The resistor Rk is
added to the circuit as shown if the cathode
impedance of the tube used is lower than the
www.americanradiohistory.com
HANDBOOK
Feedback
characteristic impedance of the cable. If the
output impedance of the stage is higher than
the cable impedance a resistance of appropriate value is sometimes placed in parallel
with the input end of the cable. The values
of Cd and Rd should be chosen with the same
considerations in mind as mentioned in the
discussion of the circuit of figure 26C above.
INPUT SIGNAL ES
The
may
6 -15
Feedback Amplifiers
It is possible to modify the characteristics
of an amplifier by feeding back a portion of
the output to the input. All components, circuits and tubes included between the point
where the feedback is taken off and the point
where the feedback energy is inserted are
said to be included within the feedback loop.
An amplifier containing a feedback loop is
said to be a feedback amplifier. One stage or
any number of stages may be included within
the feedback loop. However, the difficulty of
obtaining proper operation of a feedback amplifier increases with the bandwidth of the
amplifier, and with the number of stages and
circuit elements included within the feedback
loop.
The gain and phase
shift of any amplifier
are functions of frequency. For any amplifier containing a feedback loop to be completely stable the gain of
such an amplifier, as measured from the input
back to the point where the feedback circuit
connects to the input, must be less than one
Gain and Phase -shift
in Feedback Amplifiers
uTPUT
E
A
VOLTAGE AMPLIFICATION WITH FEEDBACK
1
cathode follower
conveniently be
used asa method of coupling r -f or i -f energy between two units separated a considerable distance. In such an
application a coaxial cable should be used to
carry the r -f or i -f energy. One such application would be for carrying the output of a v -f -o
to a transmitter located a considerable distance from the operating position. Another
application would be where it is desired to
feed a single -sideband demodulator, an FM
adaptor, or another accessory with intermediate frequency signal from a communications receiver. A tube such as a 6CB6 connected in a manner such as is shown in figure
26D would be adequate for the i -f amplifier
coupler, while a 6L6 or a 6AG7 could be used
in the output stage of a v -f -o as a cathode
follower to feed the coaxial line which carries
the v -f-o signal from the control unit to the
transmitter proper.
AMPLIFIER
GAIN= A
FEEDBACK OR B PATH
A
The Cathode -Follower
in R -F Stages
129
Amplifiers
FEEDBACK IN DECIBELS
=
GAIN IN ABSENCE
=
B
8
-A
OF
B
FEEDBACK
FRACTION OF OUTPUT VOLTAGE FED BACK
=
NEGATIVE FOR NEGATIVE FEEDBACK
IS
20 LOG
(1
-A8)
- 20 L0G MID FRED- GAIN WITHOUT FEEDBACK
MIDFREO. GAIN WITH FEEDBACK
DISTORTION WITH FEEDBACK
RD
_
DISTORTION WITHOUT FEEDBACK
(1
8)
-A
RN
_
1
-Aa
(1+
-)
WHERE
RD =OUTPUT IMPEDANCE OF AMPLIFIER WITH FEEDBACK
RNA OUTPUT IMPEDANCE
RL
=
OF
AMPLIFIER WITHOUT FEEDBACK
LOAD IMPEDANCE INTO WHICH AMPLIFIER OPERATES
Figure 28
FEEDBACK AMPLIFIER RELATIONSHIPS
at the frequency where the feedback voltage
is in phase with the input voltage of the amplifier. If the gain is equal to or more than
one at the frequency where the feedback voltage is in phase with the input the amplifier
will oscillate. This fact imposes a limitation
upon the amount of feedback which may be
employed in an amplifier which is to remain
stable. If the reader is desirous of designing
amplifiers in which a large amount of feedback is to be employed he is referred to a
book on the subject by H. W. Bode.
Feedback may be either negative
positive, and the feedback voltage may be proportional either to
output voltage or output current. The most
commonly used type of feedback with a -f or
video amplifiers is negative feedback proportional to output voltage. Figure 28 gives
the general operating conditions for feedback
amplifiers. Note that the reduction in distortion is proportional to the reduction in gain of
the amplifier, also that the reduction in the
output impedance of the amplifier is somewhat
greater than the reduction in the gain by an
amount which is a function of the ratio of the
Types of
Feedback
or
H. W. Bode,
fier
Dsign,'
and Feedback AmpliVan Nostrand Co., 250 Fourth Ave.,
"Network Analysis
D.
New York 3, N. Y.
www.americanradiohistory.com
130
Vacuum
THE
Amplifiers
Tube
RADIO
Figure 29 illustrates a very simple and effective application of negative voltage feedback to an output pentode or tetrode amplifier
stage. The reduction in hum and distortion
may amount to 15 to 20 db. The reduction in
the effective plate impedance of the stage will
be by a factor of 20 to 100 dependent upon the
operating conditions. The circuit is commonly
used in commercial equipment with tubes such
as the 6SJ7 for VI and the 6V6 or 6L6 for V2.
O° FEEDBACK
20 LOG
e
a
20L04
GAIN OP BOTH sTNGES
R2
* R. RZ(G..V2
Ra
+RR
I
¡l
I
=
RO)
(VOLTAGE CAIN
[
Goo,
(
6 -16
OrV2))
R2
)
::.!-:1)
RN
;
R(G..vz Ro)
111
WHERE.
RNR°
RD
=
R2
OUTPUT
R, %RD
R, +R2
Rz
GN.z Ro
RCrLECTt0 LOAD IMPEDANCE
.NEDPNCE
RN
=
-
RN
ON
V2
(USUALLY ABOUT S00 R)
PEED°ACN RESISTOR
R2
iRZ+RN(GwV2RO))(.+
PLATE IMPEDANCE
R
)
Vacuum -Tube Voltmeters
The vacuum -tube voltmeter may be considered
to be a vacuum -tube detector in which the
rectified d -c current is used as an indication
of the magnitude of the applied alternating
voltage. The vacuum tube voltmeter (v.t.v.m.)
consumes little or no power and it may be
calibrated at 60 cycles and used at audio or
radio frequencies with little change in the
calibration.
or V2
si mple v.t.v.m. is
shown in figure 30.
The plate load may be
a mechanical device, such as a relay or a
meter, or the output voltage may be developed
across a resistor and used for various control purposes. The tube is biased by Ec and
a fixed value of plate current flows, causing
a fixed voltage drop across the plate load
resistor, Rp. When a positive d -c voltage is
applied to the input terminals it cancels part
of the negative grid bias, making the grid
more positive with respect to the cathode.
This grid voltage change permits a greater
amount of plate current to flow, and develops
a greater voltage drop across the plate load
resistor. A negative input voltage would decrease the plate current and decrease the
voltage drop across Rp, The varying voltage
drop across Rp may be employed as a control
voltage for relays or other devices. When it is
desired to measure various voltages, a voltage
Basic
Figure 29
SHUNT FEEDBACK CIRCUIT
FOR PENTODES OR TETRODES
This circuit requires only the addition of
one resistor, R2, to the normal circuit for
such an application. The plate impedance
and distortion Introduced by the output
stage are materially reduced.
output impedance of the amplifier without
feedback to the load impedance. The reduction
in noise and hum in
those
stages included
within the feedback loop is proportional to the
reduction in gain. However, due to the reduction in gain of the output section of the amplifier somewhat increased gain is required of
the stages preceding the stages included within the feedback loop. Therefore the noise and
hum output of the entire amplifier may or may
not be reduced dependent upon the relative
contributions of the first part and the latter
part of the amplifier to hum and noise. If most
of the noise and hum is coming from the stages
included within the feedback loop the undesired signals will be reduced in the output
from the complete amplifier. It is most frequently true in conventional amplifiers that
the hum and distortion come from the latter
stages, hence these will be reduced by feedback, but thermal agitation and microphonic
noise come from the first stage and wilt not
be reduced but may be increased by feedback
unless the feedback loop includes the first
stage of the amplifier.
D -C Vacuum Tube Voltmeter
A
Figure 30
SIMPLE VACUUM TUBE
VOLTMETER
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Vacuum Tube Voltmeters
HANDBOOK
131
ZERO- ADJUST
Figure 31
D -C
VACUUM TUBE VOLTMETER
Figure 32
BRIDGE -TYPE VACUUM TUBE
VOLTMETER
range switch (figure 31) may precede the v.t.
v.m. The voltage to be measured is applied to
voltage divider, R1, R2, R3, by means of the
"voltage range" switch. Resistor R4 is used
to protect the meter from excessive input
voltage to the v.t.v.m. In the plate circuit of
the tube an additional battery and a variable
resistor ( "zero adjustment ") are used to
balance out the meter reading of the normal
plate current of the tube. The zero adjustment
potentiometer can be so adjusted that the
meter M reads zero current with no input voltage to the v.t.v.m. When a d -c input voltage
is applied to the circuit, current flows through
the meter, and the meter reading is proportional to the applied d -c voltage.
The Bridge -type
V.T.V.M.
Another important use
of a d-c amplifier is to
show the exact point of
voltages. This is
done by means of a bridge circuit with two
d -c amplifiers serving as two legs of the
bridge (figure 32). With no input signal, and
with matched triodes, no current will be read
on meter M, since the IR drops across Ri and
R2 are identical. When a signal is applied to
one tube, the IR drops in the plate circuits
become unbalanced, and meter M indicates
the unbalance. In the same way, two d -c voltages may be compared if they are applied to
the two input circuits. When the voltages are
equal, the bridge is balanced and no current
flows through the meter. If one voltage changes,
the bridge becomes unbalanced and indication
of this will be noted by a reading of the meter.
balance between two
d -c
For the purpose of analysis,
the operation of a modern
v.t.v.m. will be described. The lleatbkit V -7A
isa fit instrument for such adescription, since
it is able to measure positive or negative d -c
potentials, a -c r -m -s values, peak -to-peak
values, and resistance. The circuit of this
unit is shown in figure 33. A sensitive 200 d -c
A Modern VTVM
microammeter is placed in the cathode circuit
of a 12AU7 twin triode. The zero adjust control
sets up a balance between the twosections of
the triode such that with zero input voltage
applied to the first grid, the voltage drop across
each portion of the zero adjust control is the
same. Under this condition of balance the
meter will read zero. When a voltage is applied
to the first grid, the balance in the cathode
circuits is upset and the meter indicates the
degree of unbalance. The relationship between
the applied voltage on the first grid and the
meter current is linear and therefore the meter
can be calibrated with a linear scale. Since
the tube is limited in the amount of current
it can draw, the meter movement is elec-
tronically protected.
The maximum test voltage applied to the
12AU7 tube is about 3 volts. Higher applied
voltages are reduced by a voltage divider
which has a total resistance of about 10
megohms. An additional resistance of 1- megohm
is located in the d -c test prod, thereby permitting measurements to be made in high impedance circuits with minimum disturbance.
The rectifier portion of the v.t.v.m. is shown
in figure 34. When a -c measurements are desired, a 6AL5 double diode is used as a full
wave rectifier to provide a d -c voltage pro portionalto the applied a-c voltage. This d -c
voltage is applied through the voltage divider
string to the 12AU7 tube causing the meter to
indicate in the manner previously described.
The a -c voltage scales of the meter are calibrated in both RMS and peak-to -peak values.
In the 1.5, 5, 15, 50, and 150 volt positions
of the range switch, the full a -c voltage being
measured is applied to the input of the 6AL5
full wave rectifier. On the 500 and 1500 volt
positions of the range switch, a divider network reduces the applied voltage in order to
limit the voltage input to the 6AL5 to a safe
recommended level.
www.americanradiohistory.com
132
THE
HEATHKIT PEAKTO -PEAK
MODEL V -7A
Figure
VTVM
33
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RADIO
Vacuum Tube Voltmeters
HANDBOOK
02
A -C
150v
133
SNIELDED PROBE CASE
6AL5
INPUT 0---I
22
OM5
MAX
MEG
ff
VTVM
INPUT
JACK
TO
DC
COAA,L LINE
A 7
MEG
OOSi
=PROBE
TIP
Ce-70 S
Figure
34
Figure
FULL -WAVE RECTIFIER
FOR V.T.V.M.
35
-F PROBE SUITABLE
FOR USE IN IKC -100 MC
R
RANGE
The a -c calibrate control (figure 33) is used
to obtain the proper meter deflection for the
applied a -c voltage. Vacuum tubes develop a
contact potential between tube elements. Such
contact potential developed in the diode would
cause a slight voltage to be present at all
times. This voltage is cancelled out by proper
application of a bucking voltage. The amount
of bucking voltage is controlled by the a -c
balance control. This eliminates zero shift
of the meter when switching from a -c to d -c
readings.
For resistance measurements, a 1.5 volt
battery is connected through a string of multipliers and the external resistance to be measured, thus forming a voltage divider across
the battery, and a resultant portion of the
battery voltage is applied to the 12AU7 twin
triode. The meter scale is calibrated in resistance (ohms) for this function.
Test Probes
Auxiliary test probes may
be used with the v.t.v.m.
to extend the operating range, or to measure
radio frequencies with high accuracy. Shown
in figure 35 is a radio frequency probe which
provides linear response to over 100 megacycles. A crystal diode is used as a rectifier,
and d-c isolation is provided by a .005 uufd
capacitor. The components of the detector are
mounted within a shield at the end of a length
of coaxial line, which terminates in the d-c
input jack of the v.t.v.m. The readings obtained are RM1S, and should be multiplied by
1.414 to convert to peak readings.
www.americanradiohistory.com
CHAPTER SEVEN
High
Fidelity Techniques
The art and science of the reproduction of
sound has steadily advanced, following the
major audio developments of the last decade.
Public acceptance of home music reproduction
on a "high fidelity" basis probably dates from
the summer of 1948 when the Columbia L -P
microgroove recording techniques were introduced.
The term high fidelity refers to the reproduction of sound in which the different distortions of the electronic system are held below
limits which are audible to the majority of
listeners. The actual determination, therefore,
of the degree of fidelity of a music system is
largely psychological as it is dependent upon
the ear and temperament of the listener. By and
large, a rough area of agreement exists as to
what boundaries establish a "hi -fi" system. To
enumerate these boundaries it is first necessary
to examine sound itself.
As shown in figure 1, the sound wave of
the fork has frequency, period, and pitch. The
frequency is a measure of the number of vibrations per second of the sound. A fork tuned
to produce 261 vibrations per second is tuned
to the musical note of middle -C. It is of interest to note that any object vibrating, moving,
or alternating 261 times per second will produce a sound having the pitch of middle -C.
The pitch of a sound is that property which is
determined by the frequency of vibration of
the source, and not by the source itself. Thus
an electric dynamo producing 261 c.p.s. will
have a hum -pitch of middle -C, as will a siren,
a gasoline engine, or other object having the
same period of oscillation.
))111I)
7 -1 The Nature of Sound
))'
III
Experiments with a simple tuning fork in
the seventeenth century led to the discovery
that sound consists of a series of condensations
and rarefactions of the air brought about by
movement of air molecules. The vibrations of
the prongs of the fork are communicated to
the surrounding air, which in turn transmits
the agitation to the ear drums, with the result
that we hear a sound. The vibrating fork produces a sound of extreme regularity, and this
regularity is the essence of music, as opposed to
noise which has no such regularity.
TUNING FORK
Figure
1
VIBRATION OF TUNING FORK PRODUCES A SERIES OF CONDENSATIONS
AND RAREFACTIONS OF AIR MOLECULES. THE DISPLACEMENT OF AIR
MOLECULES CHANGES CONTINUALLY
WITH RESPECT TO TIME, CREATING
A SINE WAVE OF MOTION OF THE
DENSITY VARIATIONS.
134
www.americanradiohistory.com
Nature of Sound
FREQUENCY (CYCLES
NOTE
D
C
EQUAL-
TEMPERED 261.0
E
F
PER SECOND)
G
A
B
-I
W
I
H
C'
293.7 329.6 349.2 392.0 440.0 493.9 523.21
SCALE
I
Figure 2
EQUAL- TEMPERED SCALE
CONTAINS TWELVE INTERVALS, EACH OF
WHICH IS 1.06 TIMES THE FREQUENCY OF THE NEXT LOWEST. THE HALFINCLUDE
THE
INTERVALS
TONE
ABOVE NOTES PLUS FIVE ADDITIONAL NOTES: 277.2, 311.1, 370, 415.3,
466.2 REPRESENTED BY THE BLACK
KEYS OF THE PIANO.
THE
135
C
7
T/ME
-J
a.
2
Figure 3
THE COMPLEX SOUND OF A MUSICAL
INSTRUMENT IS A COMBINATION OF
SIMPLE SINE -WAVE SOUNDS, CALLED
HARMONICS. THE SOUND OF LOWEST
FREQUENCY IS TERMED THE FUNDAMENTAL. THE COMPLEX VIBRATION
OF A CLARINET REED PRODUCES A
SOUND SUCH AS SHOWN ABOVE.
The Musical
The musical scale is composed
of notes or sounds of various
frequencies that bear a pleasing
aural relationship to one another. Certain combinations of notes are harmonious to the ear
if their frequencies can be expressed by the
simple ratios of 1:2, 2:3, 3:4, and 4:5. Notes
differing by a ratio of 1:2 are said to be separated by an octave.
The frequency interval represented by an
octave is divided into smaller intervals, forming the musical scales. Many types of scales
have been proposed and used, but the scale of
the piano has dominated western music for the
last hundred or so years. Adapted by J. S.
Bach, the equal- tempered scale ( figure 2) has
twelve notes, each differing from the next by
the ratio 1:1.06. The reference frequency, or
American Standard Pitch is A, or 440.0 cycles.
Scale
Harmonics and
Overtones
The complex sounds pro duced by a violin or a wind
instrument bear little resemblance to the simple sound wave of the tuning
fork. A note of a clarinet, for example (when
viewed on an oscilloscope) resembles figure 3.
Vocal sounds are even more complex than this.
In 1805 Joseph Fourier advanced his monumental theorem that made possible a mathematical analysis of all musical sounds by showing that even the most complex sounds are
made up of fundamental vibrations plus harmonics, or overtones. The tonal qualities of
any musical note may be expressed in terms of
the amplitude and phase relationship between
the overtones of the note.
To produce overtones, the sound source must
be vibrating in a complex manner, such as is
shown in figure 3. The resulting vibration is
a combination of simple vibrations, producing
a rich tone having fundamental, the octave
tone, and the higher overtones. Any sound
-
-
no matter how complex
can be analyzed
into pure tones, and can be reproduced by a
group of sources of pure tones. The number
and degree of the various harmonics of a tone
and their phase relationship determine the
quality of the tone.
For reproduction of the highest quality, these
overtones must be faithfully reproduced. A musical note of 523 cycles may be rich in twentieth order overtones. To reproduce the original quality of the note, the audio system must
be capable of passing overtone frequencies of
the order of 11,000 cycles. Notes of higher
fundamental frequency demand that the audio
system be capable of good reproduction up to
the maximum response limit of the human
ear, in the region of 15,000 cycles.
Reproduction
Many factors enter into the
problem of high quality audio
reproduction. Most important
of these factors influence the overall design of
the music system. These are:
Restricted frequency range.
Nonlinear distortions.
Limitations
1-
23-Transient distortion.
4- Nonlinear frequency response.
-Phase distortion.
6- Noise, "wow ", and "flutter ".
5
A restricted frequency range of reproduction
will tend to make the music sound "tinny"
and unrealistic. The fundamental frequency
range covered by the various musical instruments and the human voice lies between 15
cycles and 9,000 cycles. Overtones of the instruments and the voice extend the upper
audible limit of the music range to 15,000
cycles or so. In order to fully reproduce the
musical tones falling within this range of frequencies the music system must be capable of
flawlessly reproducing all frequencies within
the range without discrimination.
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THE RADIO
High Fidelity Techniques
136
BASIC LIMITS FOR HIGH FIDELITY AND
"GOOD QUALITY+ REPRODUCTION
TYPE OF
DISTORTION
RESTRICTED
FREQUENCY
RANGE
LIMIT
HIGH FIDELITY
20 -IS000 CPS
INTER MODULATION
DISTORTION AT
FULL OUTPUT
4
%
2
%
HARMONIC
DISTORTION AT
FULL OUTPUT
WOW"
HUM AND NOISE
"GOODY REPRODUCTI I
SO
-10000
10 W.
S
0.1%
-70
DB BELOW
FULL OUTPUT
CPS
%
1W.
-50
OB BELOW
FULL OUTPUT
Figure 4
Nonlinear qualities such as harmonic and
intermodulation (IM) distortion are extremely
objectionable and are created when the output
of the music system is not exactly proportional
to the input signal. Nonlinearity of any part
of the system produces spurious harmonic frequencies, which in turn lead to unwanted beats
and resonances. The combination of harmonic
frequencies and intermodulation products produce discordant tones which are disagreeable
to the ears.
The degree of intermodulation may be measured by applying two tones f1 and f: of known
amplitude to the input of the amplifier under
test. The relative amplitude of the difference
tone (f2-f1) is considered a measure of the
intermodulation distortion. Values of the order
of 4% IM or less define a high fidelity music
amplifier.
Response of the music system to rapid transient changes is extremely important. Transient peaks cause overloading and shock- excitation of resonant circuits, leaving a "hang- over"
effect that masks the clarity of the sound. A
system having poor transient response will not
sound natural to the ear, even though the distortion factors are acceptably low.
Linear frequency response and good power
handling capability over the complete audio
range go hand in hand. The response should
be smooth, with no humps or dips in the curve
over the entire frequency range. This requirement is particularly important in the electromechanical components of the music system,
such as the phonograph pickup and the loudspeaker.
Phase distortion is the change of phase angle between the fundamental and harmonic
frequencies of a complex tone. The output
wave envelope therefore is different from the
envelope of the input wave. In general, phase
distortion is difficult to hear in sounds having
complex waveforms and may be considered to
be sufficiently low in value if the IM figure of
the amplifier is acceptable.
Noise and distortions introduced into the
program material by the music system must
be kept to a minimum as they are particularly
noticeable. Record scratch, turntable "rumble",
and "flutter" can mar an otherwise high quality system. Inexpensive phonograph motors do
not run at constant speed, and the slight variations in speed impart a variation in pitch
( wow) to the music which can easily be heard.
Vibration of the motor may be detected by the
pickup arm, superimposing a low frequency
rumble on the music.
The various distortions that appear in a
music system are summarized in figure 4, together with suggested limits within which the
system may truly be termed "high fidelity."
7 -2 The Phonograph
The modern phonograph record is a thin
disc made of vinylite or shellac material. Disc
rotation speeds of 78.3, 33 1/3, and 45 r.p.m.
are in use, with the older 78.3 r.p.m. speed
gradually being replaced by the lower speeds.
A speed of 16 2/3 r.p.m. is used for special
"talking book" recordings. A continuous groove
is cut in the record by the stylus of the recording machine, spiralling inward towards the
center of the record. Amplitude variations in
this groove proportional to the sound being
recorded constitute the means of placing the
intelligence upon the surface of the record.
The old 78.3 r.p.m. recordings were cut approximately 100 grooves per inch, while the
newer "micro- groove" recordings are cut approximately 250 grooves per inch. Care must
be taken to see that the amplitude excursions
of one groove do not fall into the adjoining
groove. The groove excursions may be controlled by the system of recording, and by
equalization of the recording equipment.
The early commercial phonograph records were cut with a
mechanical- acoustic system that
produced a constant velocity characteristic with
the amplitude of cut increasing as the recorded
frequency decreased ( figure 5A) . When the
recording technique became advanced enough
to reproduce low audio frequencies, it was
necessary to reduce the amplitude of the lower
frequencies to prevent overcutting the record.
A crossover point near 500 cycles was chosen,
Recording
Techniques
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HANDBOOK
High Fidelity Amplifier
CONSTANT AMPLITUDE,
137
CROSSOVER
FREQUENCY
CONSTANT
VELOCITY
SURFACE NO /SE
SURFACE NO /SE
L__ 4
f
4
SO
100
200
-1-
1
4
500
1000
FREQUENCY
2000
5000
50
100
I
500
FREQUENCY
(cvs)
1000
\
2000
5000
'
(CPS)
CONSTANT AMPLITUDE BELOW CROSSOVER
FREQUENCY. CONSTANT VELOCITY ABOVE
CROSSOVER FREQUENCY
CONSTANT VELOCITY RECORDING
V--
t
200
CROSSOVER
FREQUENCY
SURFACE NO
SURFACE NOISE
50
100
200
1000
500
2000
5000
SO
200
100
500
FREQUENCY
FREQUENCY (,P-)
CONSTANT AMPLITUDE BELOW CROSSOVER
FREQUENCY, NIGH FREQUENCY PRE -EMPHASIS
ABOVE CROSSOVER FREQUENCY
1000
/SE
2000
solo°
(cps)
RESPONSE OF RECORD OF SC PLAYED ON
PROPERLY COMPENSATED EQUIPMENT
Figure 5
MODERN PHONOGRAPH RECORD EMPLOYS CONSTANT AMPLITUDE CUT
BELOW CROSSOVER POINT AND HIGH
FREQUENCY PRE- EMPHASIS (BOOST)
ABOVE CROSSOVER FREQUENCY.
"RIAA" PLAYBACK
10
20
40
70100
300 500
CURVE
1000
3000
10000 30000
FREQUENCY. CPS
The most popular types
of pickup cartridges in
use today are the high
impedance crystal unit, and the low impedance variable reluctance cartridge. The
crystal pickup consists of a Rochelle salt
element which is warped by the action of
the phonograph needle, producing an electrical impulse whose frequency and amplitude are proportional to the modulation of
the record groove. One of the new "transducer"
crystal cartridges is shown in figure 6. When
working into a high impedance load, the output of a high quality crystal pickup is of the
The Phonograph
Pickup
and a constant amplitude groove was cut below
this frequency ( figure 5B). This system does
not reproduce the higher audio notes, since the
recording level rapidly drops into the surface
noise level of the record as the cutting frequency is raised. The modern record employs
pre-emphasis of the higher frequencies to boost
them out of the noise level of the record
(figure 5C) . When such a record is played
back on properly compensated equipment, the
audio level will remain well above the background noise level, as shown in figure 5D.
www.americanradiohistory.com
138
THE RADIO
High Fidelity Techniques
Figure 8
Figure 6
NEW CRYSTAL "TRANSDUCER"
CARTRIDGE PROVIDES HIGH FIDELITY OUTPUT AT RELATIVELY
HIGH LEVEL
order of one -half volt or so. Inexpensive crystal
units used in 78 r.p.m. record changers and
ac -dc phonographs may have as much as two
or three volts peak output. The frequency response of a typical high quality crystal pickup
is shown in figure 7.
The variable reluctance pickup is shown in
figure 8. The reluctance of the air gap in a
magnetic circuit is changed by the movement
of the phonograph needle, creating a variable
voltage in a small coil coupled to the magnetic
lines of force of the circuit. The output impedance of the reluctance cartridge is of the
order of a few hundred ohms, and the output
is approximately 10 millivolts.
For optimum performance, an equalized preamplifier stage is usually employed with the
reluctance pickup. The circuit of a suitable unit
is shown in figure 9. Equalization is provided
by R5, R,, and C., with a low frequency crossover at about 500 cycles. Total equalization is
15 db. High frequency response may be limited
by reducing the value of R. to 5,000
15,000
ohms.
The standard
records has a tip
the microgroove
has a tip radius
-
pickup stylus for 78 r.p.m.
radius of .0025 inch, whereas
(33 1/3 and 45 r.p.m.) stylus
of .001 inch. Many pickups,
"RELUCTANCE" CARTRIDGE
STANDARD PICK -UP FOR
IS
MUSIC SYSTEM.
Low stylus pressure of four grams insures
minimum record wear. Dual stylus is used
having two needle tip diameters for long
playing and 78 R.P.M. recordings.
therefore, are designed to have interchangeable
cartridges or needles to accomodate the different groove widths.
7 -3
The High Fidelity Amplifier
A block diagram of a typical high fidelity
system is shown in figure 10. A preamplifier
is used to boost the output level of the phonograph pickup, and to permit adjustment of input selection, volume, record compensation,
and tone control. The preamplifier may be
mounted directly at the phonograph turntable
position, permitting the larger power amplifier
to be placed in an out of the way position.
The power amplifier is designed to operate
from an input signal of a volt or so derived
from the preamplifier, and to build this signal
to the desired power level with a minimum
amount of distortion. Maximum power output
levels of ten to twenty watts are common for
home music systems.
The power supply provides the smoothed,
d -c voltages necessary for operation of the preamplifier and power amplifier, and also the
6SC7
SNORT
a
w
OUTPUT
LEADS TO
RELUCTANCE
CARTRIDGE
+10
Rts
+5Rz
33M
Z 0
a -5
20
r
e
R3
50
too
Re
200
500
FREQUENCY
1000 2000 5000
68K
10000
C4 C5
T-
e
Rs
33K
e+ 100 v
HIGH PHONOGRAPH
QUALITY
CRYSTAL
CARTRIDGE. (ELECTROVOICE
56 -DS
POWER POINT TRANSDUCER)
RESPONSE
,33K
(cps)
Figure 7
FREQUENCY
Ce
.0
RT
66
=
TC3
_ .01
-10
Lai
¢
66K
4
OF
Figure 9
PREAMPLIFIER SUITABLE FOR USE WITH
LOW LEVEL RELUCTANCE CARTRIDGE.
www.americanradiohistory.com
i
HANDBOOK
!PHONOGRAPH
High Fidelity Amplifier
139
+20
RI
f
1
M
SPEAKER
ENCLOSURE
POWER
SUPPLY
-20
20
100
SO
200
SOO
FREQUENCY
I.
Figure 10
BLOCK DIAGRAM OF HIGH FIDELITY
MUSIC SYSTEM.
1000 2000
N
TREBLE
5000 10000
(CeS)
Figure 12
FREQUENCY RESPONSE CURVES FOR
THE BASS AND TREBLE BOOST
AND ATTENUATION CIRCUITS
OF FIGURE 11.
TREBLE BOOST
AND ATTENUATION
BASS BOOST
AND ATTENUATION
1
SPONSE
LOUD
SPEAKER
I
v
-t
I
POWER
PREAMPLIFIER+ AMPLIFIER
S
TREBLE BOOST
BASS BOOST
TWEETERI
INPUT
RI
eóosr
C,
Rz
80ó$r
OUTPUT
OUTPUT
ATTENUATE
ArrENUATE
C2
Rn
EQUIVALENT CIRCUITS
EQUIVALENT CIRCUITS
ATTENUATE
ATTENUATE
BOOST
IN.
Ri
OUT.
OUT.
C2
Rz
SIMPLE
Equalizer
networks are employed in high fidelity equipment to 1)- tailor the response curve of the system to obtain the correct
compensate
overall frequency response, 2)
for inherent faults in the program material,
merely to satisfy the hearing preference
3
of the listener. The usual compensation networks are combinations of RC and RL networks that provide a gradual attenuation over
a given frequency range. The basic RC networks suitable for equlizer service are shown
in figure 11. Shunt capacitance is employed
for high frequency attenuation, and series
capacitance is used for low frequency attenuation. A combination of these simple a -c
voltage dividers may be used to provide almost
any response, as shown in figure 12. It is
common practice to place equalizers between
two vacuum tubes in the low level stages of
the preamplifier, as shown in figure 13. Bass
and treble boost and attenuation of the order
Tone
Compensaton
-to
C1
our
hold its own in the race for true fidelity.
Speaker efficiency runs from about 10% for
cone units to nearly 40% for high frequency
tweeters. The frequency response of any speaker is a function of the design and construction
of the speaker enclosure or cabinet that mounts
the reproducer.
O
Figure 11
CIRCUITS MAY BE
R -C
USED FOR BASS AND TREBLE
BOOST OR ATTENUATION.
filament voltages (usually a -c) for the heaters
of the various amplifier tubes.
The loudspeaker is a device which couples
the electrical energy of the high fidelity system to the human ear and usually limits the
overall fidelity of the complete system. Great
advances in speaker design have been made in
the past years, permitting the loudspeaker to
)-or
+1zAx7
.04
E. OUTPUT
220
INPUT
Figure 13
AND TREBLE LEVEL
CONTROLS, AS
EMPLOYED IN THE
HEATHKIT WA -P2
PREAMPLIFIER.
BASS
B+
BASS CONTROL
www.americanradiohistory.com
TREBLE CONTROL
K
140
THE RADIO
High Fidelity Techniques
2N190
2N190
IRK
2N 190
12V
1K
VARIABLE
RELUCTANCE
PICKUP
e
3 9K
OUTPUT
or;
TREBLE
BASS
Figure 14
TRANSISTORIZED HIGH- FIDELITY PREAMPLIFIER FOR USE WITH
RELUCTANCE PHONOGRAPH CARTRIDGE.
f100-
Loudness
Compensaron
PAIN L EVEL
+90
+BO
+70
+60
+50
+40
+30
LOWER
LIMIT
OF
HEARING
+20
+10
0
10
20
50
100
200
500
FREQUENCY
1d00 2000 5000
10000
(CPS)
200
Figure 15
THE "FLETCHER -MUNSON" CURVE
ILLUSTRATING THE INTENSITY
RESPONSE OF THE HUMAN EAR.
of 15 db may be obtained from such a circuit.
simple transistorized preamplifier using
this type of equalizing network is shown in
figure 14.
A
The minimum threshold of
hearing and the maximum
threshold of pain vary greatly with the frequency of the sound as shown
in the Fletcher- Munson curves of figure 15.
To maintain a reasonable constant tonal balance as the intensity of the sound is changed
it is necessary to employ extra bass and treble
boost as the program level is decreased. A
simple variable loudness control is shown in
figure 16 which may be substituted for the
ordinary volume control used in most audio
equipment.
The Power
The power amplifier stage of
the music system must supply
driving power for the loudspeaker. Commercially available loudspeakers
are low impedance devices which present a
Amplifier
mVIO
o
w +5
z
o
a
O1
0
SPEAKER
-5
RESPONSE
ß:
R -10
w
aw
o")
V
14
'
O
-RESONANT FREQUENCY OF
-SPEAKER - BAFFLE COMBINATION
a
a
m_
iiI
Illi
20
RI
-R2 -R
31 THREE SECTION
POTENTIOMETER, /RC
THE FOLLOWING,
TYPE, BUILT OF
R1 - lAC P011 -135
R2 -/RC MOLT /SECT /ON M13 -137
R3 - /RC MOLT /SECT /ON M13-128
10
20
50
100
SPEAKER
200
500
FREQUENCY
Figure 16
VARIABLE LOUDNESS CONTROL FOR
USE IN LOW IMPEDANCE PLATE
CIRCUITS. MAY BE PURCHASED
AS IRC TYPE LC -1 LOUDNESS
CONTROL.
1000
2000 5000
4
10000
(CPS)
Figure 17
IMPEDANCE AND FREQUENCY
RESPONSE OF "4 -OHM" 12 -INCH
SPEAKER PROPERLY MOUNTED IN
MATCHING BAFFLE.
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Ir
20000
HANDBOOK
High Fidelity Amplifier
141
FEEDBACK RESISTOR
VJ'
Figure 18
TYPICAL TRIODE
AMPLIFIER WITH
FEEDBACK LOOP.
*
° MATCHED
varying load of two to nearly one hundred
ohms to the output stage ( figure 17) . It is
necessary to employ a high quality output
transformer to match the loudspeaker load to
the relatively high impedance plate circuit of
the power amplifier stage. In general, push pull amplifiers are employed for the output
stage since they have even harmonic cancelling
properties and permit better low frequency
response of the output transformer since there
is no d -c core saturation effect present.
To further reduce the harmonic distortion
and intermodulation inherent in the amplifier
system a negative feedback loop is placed
around one or more stages of the unit. Frequency response is thereby improved, and the
output impedance of the amplifier is sharply
reduced, providing a very low source impedance for the loudspeaker.
PAIR RESISTORS
Shown in figure 18 is a basic push -pull
triode amplifier, using inverse feedback around
the power output and driver stage. A simple
triode inverter is used to provide 180- degree
phase reversal to drive the grid circuit of the
power amplifier stage. Maximum undistorted
power output of this amplifier is about 8 watts.
A modification of the basic triode amplifier
is the popular Williamson circuit (figure 19)
developed in England in 1947. This circuit
rapidly became the "standard of comparison"
in a few short years. Pentode power tubes are
connected as triodes for the output stage, and
negative feedback is taken from the secondary
of the output transformer to the cathode of
the input stage. Only the most linear portion
of the tube characteristic curve is used. Although that portion has been extended by
higher than normal plate supply voltage, it
FEEDBACK RESISTOR
SK-ISK
6SN7GT
65N7GT
807
0.25
INPUT
OUTPUT
22K
NJ
30KK{
= 1
10 20
4-400V.
AT 140 MA
Figure 19
U. S. VERSION OF BRITISH
OUTPUT AT
LESS
THAN
"WILLIAMSON" AMPLIFIER PROVIDES 10 WATTS POWER
2 °o
INTERMODULATION DISTORTION. 6SN7 STAGE
DIRECT COUPLING.
www.americanradiohistory.com
USES
142
THE RADIO
High Fidelity Techniques
807/5881
TO
FEEDBACK
CIRCUIT
0
25
FROM
05N7GT
PHASE
OUTPUT
INVERTER
0.25
807/5881
NOTE, P/N CONNECT IONS ARE
POR 807 TUBES
f00V
Figure 20
"ULTRA- LINEAR" CONFIGURATION OF WILLIAMSON AMPLIFIER DOUBLES POWER OUTPUT, AND REDUCES IM LEVEL. SCREEN TAPS ON OUTPUT TRANSFORMER PERMIT
"SEMI -TETRODE" OPERATION.
is only a fraction of the curve normally used
in amplifiers. Thus a comparatively low output
power level is obtained with tubes capable of
much more efficient operation under less
stringent requirements. With 400 volts applied to the output stage, a power output of 10
watts may be obained wtih less than 2% inter modulation distortion.
A recent variation of the Williamson circuit involves the use of a tapped output transformer. The screen grids of the push -pull amplifier stage are connected to the primary taps,
allowing operating efficiency to approach that
of the true pentode. Power output in excess
of 25 watts at less than 2% intermodulation dis-
Figure 21
"BABY HI -FI" AMPLIFIER IS DWARFED
BY 12 -INCH SPEAKER ENCLOSURE
This miniature music system is capable of excellent performance in the small home or
apartment. Preamplifier, bass and treble controls, and volume control are all incorporated
in the unit. Amplifier provides 4 watts output
at 4 IM distortion.
tortion may be obtained with this circuit
(figure 20) .
7 -4
Amplifier Construction
Wiring
Assembly and layout of high
fidelity audio amplifiers follows the general technique described for other forms of electronic equipment.
Extra care, however, must be taken to insure
that the hum level of the amplifier is extremely low. A good hi -fi system has excellent response in the 60 cycle region, and even a
minute quantity of induced a -c voltage will be
disagreeably audible in the loudspeaker. Spurious eddy currents produced in the chassis by
the power transformer are usually responsible
for input stage hum.
To insure the lowest hum level, the power
transformer should be of the "upright" type
instead of the "half-shell" type which can
couple minute voltages from the windings to
a steel chassis. In addition, part of the windings
of the half -shell type project below the chassis
where they are exposed to the input wiring of
the amplifier. The core of the power transformer should be placed at right angles to the
core of a nearby audio transformer to reduce
spurious coupling between the two units to a
Techniques
minimum.
It is common practice in amplifier design
to employ a ground bus return system for all
audio tubes. All grounds are returned to a
single heavy bus wire, which in turn is
grounded at one point to the metal chassis.
This ground point is usually at the input jack
of the amplifier. When this system is used,
a -c chassis currents are not coupled into the
amplifying stages. This type of construction is
illustrated in the amplifiers described later in
this chapter.
www.americanradiohistory.com
HANDBOOK
Amplifier Construction
NO
FOR
RI=e.2
220
.05
PHONO INPUT
J,
K^
e n SPEAKER
6AQ5
12AU7
12AÚ7
143
470K
.05
e
Sp
MMr
W
R110
1,T2
len
SPKR
(HIGH LEVEL)
100
K
M
3
1.e
1.9K
SISOK
R
10011
OS
M
70 K
BASE
5
6AQ5
VOLUME
CONTROL
CONTROL
1,7
TREBLE
CONTROL
+203
10
450V
V.
1211
I
+210V
6X5
TI
CH1
3K 2W
2
115V
ti
IxÇ,A
=4,5
9
12AÚ7
T,- 260 -0-260
?
124Ú7
.20C111
=3
TOC1C
3
4
NOTES
ALL RESISTORS 0.3 WATT
UNLESS OTHERWISE SPECIFIED
2. ALL CAPACITOR VALUES IN MF
UNLESS OTHERWISE SPECIFIED
3. RESISTORS MARKED A ARE
MATCHED PAIRS
1.
=
4
6.405
RAO5
Figure 22
SCHEMATIC, "BABY HI -FI" AMPLIFIER
CH,-1.5 henry at 200 ma. Chicago Standard
6.3 volts at 4.0
volts at 90 ma.,
amp., upright mounting. Chicago -Standard
PC -8420.
T,-10 K, CT. to 8, 16 ohms. Peerless (Altec)
S -510F.
Care should be taken to reduce the capacitance to the chassis of high impedance circuits,
or the high frequency response of the unit will
suffer. Shielded "bath -tub" type capacitors
should not be used for interstage coupling capacitors. Tubular paper capacitors are satisfactory. These should be spaced well away from
the chassis.
It is a poor idea to employ the chassis as a
common filament return, especially for low
level audio stages. The filament center-tap of
the power transformer should be grounded,
and twisted filament wires run to each tube
socket. High impedance audio components and
wiring should be kept clear of the filament
lines, which may even be shielded in the vicinity of the input stage. In some instances, the
filament center tap may be taken from the arm
of a low resistance, wirewound potentiometer
placed across the filament pins of the input
tube socket. The arm of this potentiometer is
grounded, and the setting of the control is adjusted for minimum speaker hum.
7 -5 The "Baby Hi
Fi"
A definite need exists for a compact, high
fidelity audio amplifier suitable for use in the
small home or apartment. Listening tests have
shown that an average power level of less than
C,A- B-
C- 30 -20 -10
C -2327.
Aid. 350 volt. Mallory Fp -330.7
NOTE -Feedback loop returns to 8 ohm tap on T,
when 8 ohm speaker is used.
one watt in a high efficiency speaker will provide a comfortable listening level for a small
room, and levels in excess of two or three watts
are uncomfortably loud to the ear. The "Baby
Hi -Fi" amplifier has been designed for use
in the small home, and will provide excellent
quality at a level high enough to rattle the
windows.
Designed around the new Electro -Voice miniature ceramic cartridge, the amplifier will provide over 4 watts power, measured at the secondary of the output transformer. At this level,
the distortion figure is below 1 %, and the IM
figure is 4 %. At normal listening levels, the IM
is much lower, as shown in figure 24.
The Amplifier
The schematic of the ampli fier is shown in figure 22.
Bass and treble boost controls are incorporated in the circuit, as is the
volume control. A dual purpose 12AU7 double
triode serves as a voltage amplifier with cathode degeneration. A simple voltage divider
network is used in the grid circuit to prevent
amplifier overloading when the ceramic cartridge is used. The required input signal for
maximum output is of the order of 0.3 volts.
The output level of the Electro -Voice cartridge
is approximately twice this, as shown in figure
7. The use of the high -level cartridge eliminCircuit
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144
THE RADIO
High Fidelity Techniques
Figure 23
UNDER -CHASSIS
VIEW OF
"BABY HI -FI"
Low level audio stages are
at
upper left, with components
mounted between socket pins
and potentiometer controls.
6X5 socket is at lower cen-
ter of
photo
with
filter
choke CH, at right. Feedback resistor R, is at left
of rectifier socket.
ates the necessity of high gain amplifiers required when low level magnetic pickup heads
are used. Problems of hum and distortion introduced by these extra stages are thereby
eliminated, greatly simplifying the amplifier.
The second section of the 12AU7 is used for
bass and treble boost. Simple R -C networks
are placed in the grid circuit permitting gain
boost of over 12 db at the extremities of the
response range of the amplifier.
A second 12AU7 is employed as a direct
coupled "hot-cathode" phase inverter, capacitively coupled to two 6AQ5 pentode connected
output tubes. The feedback loop is run from
the secondary of the output transformer to the
cathode of the input section of the phase inverter.
The power supply of the "Baby Hi -Fi" consists of a 6X5 -GT rectifier and a capacitor input filter. A second R -C filter section is used
to smooth the d -c voltage applied to the
12AU7 tubes. A cathode-type rectifier is used
in preference to the usual filament type to prevent voltage surges during the warm -up period
of the other cathode -type tubes.
Amplifier
The complete amplifier is
built upon a small "amplifier
foundation" chassis and cover
measuring 5 "x7 "x6" (Bud CA- 1754). Height
of the amplifier including dust cover is 6 ".
The power transformer (T,) and output transformer (Tl) are placed in the rear corners of
the chassis, with the'6X5 -GT rectifier socket
placed between them. The small filter choke
(CH,) is mounted to the wall of the chassis
and may be seen in the under -chassis photograph of figure 23. The four audio tubes are
placed in a row across the front of the chassis.
Viewed from the front, the 12AU7 tubes are
to the left, and the 6AQ5 tubes are to the right.
The three section filter capacitor (C,A, B, C)
is a chassis mounting unit, and is placed between the rectifier tube and the four audio
tubes. Since the chassis is painted, it is important that good grounding points be made at
each tube socket. The paint is cleared away
Construction
s-
t
o
a
3
EQUIVALENT SINE WAVE WATTS
Figure 24
INTERMODULATION CURVE FOR
"BABY HI -Fl" AS MEASURED ON
HEATHKIT INTERMODULATION
ANALYZER.
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HANDBOOK
"Baby Hi -Fi"
145
Figure 25
TYPICAL
INTERMODULATION
TEST OF AUDIO
AMPLIFIER.
Audio tones of two frequencies are applied to input of amplifier under testi
and amplitude of "sum" or
"difference" frequency is
measured, providing relative
inter -modulation figure.
beneath the socket bolt heads, and lock nuts
are used beneath the socket retaining nuts to
insure a good ground connection. All ground
leads of the first 12AU7 tube are returned to
the socket, whereas all grounds for the rest of
the circuit are returned to a ground lug of
filter capacitor G.
Since the input level to the amplifier is of
the order of one-half volt, the problem of
chassis ground currents and hum is not so
prevalent, as is the case with a high gain input
stage.
Phonograph -type coaxial receptacles are
mounted on the rear apron of the chassis, serving as the input and output connections. The
four panel controls (bass boost, treble boost,
volume, and a -c on) are spaced equidistant
across the front of the chassis.
Amplifier
Wiring
The filament wiring should be
done first. The center -tap of the
filament winding is grounded to
a lug of the 6X5 -GT socket ring, and the 6.3
volt leads from the transformer are attached
to pins 2 and 7 of the same socket. A twisted
pair of wires run from the rectifier socket to
the right -hand 6AQ5 socket (figure 23). The
filament leads then proceed to the next 6AQ5
socket and then to the two 12AU7 sockets in
turn.
The 12AU7 preamplifier stage is wired next.
A two terminal phenolic tie -point strip is
mounted to the rear of the chassis, holding the
12K decoupling resistor and the positive lead
of the 10 µfd., 450 -volt filter capacitor. All
B -plus leads are run to this point. Most of the
components of the bass and treble boost system
may be mounted between the tube socket terminals and the terminals of the two potentiometers. The feedback resistor R, is mounted
between the terminal of the coaxial output
connector and a phenolic tie -point strip placed
beneath an adjacent socket bolt.
When the wiring has been completed and
checked, the amplifier should be turned on,
and the various voltages compared with the
values given on the schematic. It is important
that the polarity of the feedback loop is correct.
The easiest way to reverse the feedback polarity
is to cross -connect the two plate leads of the
6AQ5 tubes. If the feedback polarization is
incorrect, the amplifier will oscillate at a supersonic frequency and the reproduced signal will
sound fuzzy to the ear. The correct connection
may be determined with the aid of an oscilloscope, as the oscillation will be easily found.
The builder might experiment with different
values of feedback resistor RI, especially if a
speaker of different impedance is employed.
Increasing the value of R1 will decrease the
degree of feedback. For an 8 -ohm speaker, Rt
should be decreased in value to maintain the
same amount of feedback.
This amplifier was used in conjunction with
General Electric S-1201A 12 -inch speaker
mounted in an Electro -Voice KD6 Aristocrat
speaker enclosure which was constructed from
a
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High Fidelity Techniques
THE RADIO
kit. The reproduction was extremely smooth,
with good balance of bass and treble.
requirements and the absence of expensive
power and audio transformers it is more economical than conventional amplifiers of similar
performance.
146
a
7 -6
A Transformerless
25 Watt Music
Amplifier
The Amplifier
The output stage of this unusual amplifier is the single
ended, push -pull type as
shown in figure 27. The quiescent current is
equal in both tubes with no d.c. current flowing through the speaker load. The absence of
an output transformer allows 40 db of feedback to be app'ied by connecting the voice
coil of the speaker directly to the cathode of
the 12ÁT7 phase- inverter driver. In addition
to its distortion reducing characteristic, the
application of feedback serves to reduce the
hum voltage which might otherwise be present.
As the gain within the feedback loop is essentially unity, an additional voltage amplifier
is used (with separate feedback) to build the
input voltage up to the voice coil level.
The power supply is a double half-wave selenium rectifier circuit developing +140 volts
and -140 volts with respect to ground. The
supply uses large filter capacitors, and no
Circuit
Because the output transformer is usually
the weakest link in both frequency response
and power output of an audio amplifier,
several methods have been used to drive loudspeakers directly from the output tubes. These
have either used non -conventional high -impedance loudspeakers, have been very inefficient,
or have had low power output capabilities.
The amplifier described in this section
drives a conventional 16 ohm loudspeaker
with normal class A amplifier efficiency, and
supplies 25 watts of low distortion output
throughout the audio range (figure 26). The
amplifier requires an input signal of approximately one volt to drive it to maximum output. The unit attains its high performance
through the use of 40 db of inverse feedback.
Because of the relatively simple power supply
Figure 26.
25 -WATT
TRANS FORMERLESS
AMPLIFIER PROVIDES ULTIMATE IN
LISTENING
PLEASURE FOR THE
"GOLDEN EAR."
Amplifier employs three
triode tubes in
single-ended push-pull
configuration for maximum fidelity. The output
6082
tubes are placed across
right end of chassis.
65N7 phase inverter is at
rear, center; and low
level stages are at the
front, center. To the left
are the power supply filter capacitors. In this
particular amplifier, the
40 ohm, 20 watt filadropping resistor
and a
iron core reactor
was used in its place
(left, rear corner of the
chassis). Across front of
chassis are (I. to r.):
power switch, input lack,
and output stage balancing potentiometer.
ment
was
small
www.americanradiohistory.com
eliminated
HANDBOOK
Transformerless Amplifier
12 AT 7
L
I
a
147
2 A T7
40
±T
+
40
150
+ 150
loo
150
47K
K
INPUT
K
6082
6SN7 -GT
0.1
*
0.1
6082
;ff-6
+
Sl
6082
56 K
5
5M
56K
4-
40
+ 150
00
100
2
6K*
5
8K
1611
OUTPUT
2
680
40
150
. IM
5
I
1.5M
M
M
1.8
10K
K
1K
4
1M
39K
00
100
56K
10
K
4
4
4
1.2
M
+250
VOLTS
+,o VOLTS
9
4,5
65147
4062 6082 6062
12AT7
6
7
7
6
6
7
11Hf--`
SR 1.5Kz
40
S6W
9
i
7
560
2.5 K
10
SI
115V1.,
NOTE:
1.
O%
40/150
12 AT7
r
n
ONE SIDE OP
70 CHASSIS.
LINE
A.
roo
fOWW
W
o
*140
óI
SR2 +aoo
GROUNDED
of
S RS
SR 4
0.5
160
ï5S
-140
VOLTS
10
A
A1°
%
T
%
REACTANCE
NETWORKS
2. RESISTORS MARRED R ARE
MATCHED PA /RS.
4.SRI.
Qx
P300500
ALL RESISTORS I -WATT UNLESS
OTHERWISE NOTED.
3. CAPACITOR VALUES
VOLTS
GIVEN IN //FD.
75 MA., I50 VOLT.
5.SRz, SR3 =500MA.,
I I
15
150 VOLT.
150
SR4
8.29
10K
27
K
12K
Figure 27.
SCHEMATIC OF 25 -WATT MUSIC AMPLIFIER
extra filtering is required. The output impedance of the supply is extremely low. To obtain higher voltage for the low level stages,
additional selenium rectifiers are used in a
voltage- adding configuration to obtain +250
and -250 volts.
about -70 volts, but for this class of service
the bias is held at -60 volts. A bias control
is provided for one set of tubes so that the
d.c. current flowing in the tubes may be
equalized, and to insure that no d.c. current
flows through the speaker voice coil.
Circuit Details
The type 6082 tube is not rated for use
with fixed bias unless a limiting resistor is
added in either the plate or the cathode circuit. Although this circuit does not use such
resistors, their omission is feasible only because the tubes are used under quiescent conditions well below maximum ratings. With
tubes of this type, it may be expected that the
average current through the voice coil will
drift with time but the presence of this un-
The complete schematic of
this amplifier is given in
figure 27. Three type 6082 double triodes are
employed in the output stage. These are 26.5
volt versions of the popular 6AS7G. These
tubes are capable of 700 milliamperes of peak
plate current per triode section at the plate
voltage employed. The choice of the 6082 is
an economy measure to allow the use of a
series heater string. These tubes cut off at
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High Fidelity Techniques
THE RADIO
balance current will generally be of little
concern. In any event, the circuit has been
designed so that the output stage can be conveniently rebalanced.
voltage than the upper group.
In the first voltage amplifier, bias is obtained from unbypassed cathode resistors since
the loss of gain can easily be tolerated. The
phase inverter- driver, however, has fixed bias
applied to the grid from the -140 volt
supply, since maximum gain is desired within
the main feedback loop.
148
The Voltage
The low level stages are all
operated Class A with conventional circuitry. A separate
driver is needed for each side of the output
circuit, as insufficient output is obtained from
the phase inverter to drive the output tubes
directly. One side of the phase inverter has a
larger load than the other, since the input to
the lower group of output tubes has the
speaker impedance in the cathode. This causes
degeneration and necessitates higher input
Amplifier
Figure 28.
UNDER -CHASSIS VIEW OF
TRANSFORMERLESS AMPLIFIER
Output tube sockets are at left, with power
supply components at right. Components of
preamplifier stages are grouped about the
center sockets, mounted between socket pins
and phenolic tie -point strips. Line fuse is
mounted on rear apron of chassis.
The Power Supply
The high current power
supply uses 300 µµfd.
filter capacitors and 5 ohm protective resistors. R -C decoupling is used to minimize hum
in the low level audio stages. As with all
"power- transformerless" equipment, care must
be taken when connecting this amplifier to
other pieces of equipment to ensure that the
grounded side of the power line is connected
to the chassis. This may be achieved by the
use of a polarized line plug, or a small isolation transformer may be employed.
The Equalizing
Circuit
As would be expected, 40
db of feedback can only be
applied within a loop having a minimum of phase shift or circuit instability will result. Since the loudspeaker
O
www.americanradiohistory.com
HANDBOOK
Transformerless Amplifier
o
149
.0
0.8
0 6
-10
0.
0.2
20
10
100
1000
10 KC
FREQUENCY
100 KC
1000 KC
Figure 29.
The balance adjustment for zero d.c. current through the speaker voice coil can be
made with a milliammeter in series with the
coil, or by measuring the voltage across the
coil with a sensitive voltmeter.
Amplifier
The amplifier is built upon
an aluminum chassis measuring 8" x 10" x 2 ". Perforated end pieces and 1/4 -inch holes drilled
around the 6082 tube sockets insure adequate
ventilation. Layout of the major components
is shown in figure 26, and placement of the
under -chassis components is shown in figure
28. As no a.c. power transformer is used,
ground currents are of small concern, and the
ground bus wiring technique need not be emConstruction
20
25
POWER OUTPUT (wnrrs)
A- Overall frequency response of amplifier
B-Distortion versus power output of amplifier
impedance becomes inductive above the audio
range it causes an increase in phase shift and
loop gain. To avoid instability an impedance
can be shunted across the voice coil to prevent
the output reactance from rising at the higher
audio frequencies. Three networks that have
been used successfully for this purpose are
shown in figure 27. The 180 ohm res for
merely limits the maximum impedance of the
output system and thus preven , excessive
feedback. The 0.5 pfd. capacitor places a low
impedance across the inductive load which is
effective at the higher audio frequencies. The
series 16 ohm resistor and 0.01 pfd. capacitor
places a resistance across the speaker at the
higher frequencies and an open circuit at the
lower frequencies. This serves to provide constant impedance and feedback over the frequency range of the amplifier.
15
10
o
0
ployed. In its place, a tinned copper wire is
run between the various chassis ground points.
Ground connections may now be made to the
socket grounding lugs, or to terminal strip
ground points. A.c. filament and power leads
are twisted wherever possible, and are run
around the outer edges of the chassis.
Point -to -point wiring technique is used,
with small capacitors and resistors mounted
to socket pins or to phenolic tie -point strips
placed near the sockets. The small silicon rectifiers are mounted to tie -point strips placed
near the upright filter capacitors.
Several of the filter capacitors do not have
their negative terminal at ground potential.
It is therefore necessary to mount the capacitor
on a phenolic plate and to slip a fiber insulating jacket over the metal shell.
Amplifier
The frequency response of
the amplifier is flat within
one db from 10 cycles to over
100 kilocycles. Since R -C coupled circuits are
used throughout, there is no serious limitation
on frequency response, and the response is
down only 4 db at 250,000 cycles. The inter stage coupling networks limit the low frequency response below 10 cycles.
Harmonic distortion and intermodulation at
full rated output are exceptionally low and virtually independent of frequency. The ability
to deliver 25 watts at 20 cycles and below
with negligible distortion is practically impossible in a transformer -type circuit of similar
mid -frequency power rating. Square wave response of the amplifier as measured between
20 cycles and 50 kilocycles is extremely good.
Performance
www.americanradiohistory.com
--IilIFte
LO-russ
T seCT,oM
LOIS
r4ss i/
011411
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VALUES
SCALE
FREQUENCY
HIGH -PASS
LOW -PASS
.00
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M1oM-491 T seCT,ON
seCT,OM
L
LOAD RESISTANCE
C
25.0
90
1000
10000
1100
9000
200
80
20.0
70
5
.5
1300
6
6
r
.7
1400
1500
.6
e
60
15.0
9
.9
10
1.0
1600
1700
W
tu
50
1600
1900
e000
Z-
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1000
6000
W
-1
u
5000 n
2000
M
4.s
W
40
10.0
V
Z 2500
<
2.0
O
4000
r
u
9.0
O 3000
Z
3.0
e.0
3000
LC
7.0
40
.0
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50
5.0
u 4000
60
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1h.
W
2
6.0
70
90
5.0
20
7.0
I
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100
9.0
10.0
4.0
--i-
-
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= 5000_4r
3
ta 1000 -- EA
Ñ
8.0
I
-5.-
15.0
C 1000
00
20.0
8000
300
30.0
150
3.0
J9000
2.5
10
10000
For both
Pi -type
1000
-
--
Courler, rrcdic e.0,9 9ni,
FILTER DESIGN
2000
n
1000
ing
CO.
CHART
and T -type Sections
connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right side scale for high -pass) with load on left -hand side of right -hand scale by means of a straight -edge.
Then read the value of L from the point where the edge intersects the left side of the center scale. Readings are in henries for frequencies in cycles per second.
To
find
L,
To find C, connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right side scale for high pass) with the load on the right -hand side of the right -hand scale. Then read the
value of C from the point where the straightedge cuts the right side of the center scale. Readings are
in microfarads for frequencies in cycles per
d.
For frequencies in kilocycles, C is expressed in thousands of micromicrofarads, L is expressed in
mlllihenries. For frequencies in megacycles, L is expressed in microhenries and C is expressed in micromlcrofarads.
For each tenfold increase In the value of load resistance multiply L by 10 and divide C by
For each ten fold decrease in frequency multiply L by 10 end multiply C by 10.
150
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IO.
CHAPTER EIGHT
Radio Frequency
Vacuum Tube Amplifiers
TUNED RF VACUUM TUBE AMPLIFIERS
Tuned r -f voltage amplifiers are used in receivers for the amplification of the incoming
r -f signal and for the amplification of intermediate frequency signals after the incoming
frequency has been converted to the intermediate frequency by the mixer stage. Signal frequency stages are normally called tuned r -f
amplifiers and intermediate -frequency stages
are called i-f amplifiers. Both tuned r -f and
i -f amplifiers are operated Class A and normally operate at signal levels from a fraction
of a microvolt to amplitudes as high as 10 to
50 volts at the plate of the last i -f stage in a
receiver.
first tuned circuit due to its equivalent coupled resistance at resonance. The noise voltage generated due to antenna radiation resistance and to equivalent tuned circuit resistance
is similar to that generated in a resistor due
to thermal agitation and is expressed by the
following equation:
En'
k
=
R =
Grid Circuit
Considerations
1f =
Since the full amplification of a receiver follows the first tuned circuit, the operating conditions existing in that circuit and in its coupling to the antenna on one side and to the
grid of the first amplifier stage on the other
are of greatest importance in determining the
signal -to -noise ratio of the receiver on weak
signals.
highest
ratio of signal -to -noise be impressed on the grid of the first
r -f amplifier tube. Attaining the optimum ratio
is a complex problem since noise will be generated in the antenna due to its equivalent
radiation resistance (this noise is in addition
to any noise of atmospheric origin) and in the
First Tuned
Circuit
It is obvious that the
4kTRAf
Where: E° = r -m -s value of noise voltage over
the interval .1f
T =
8 -1
=
Boltzman's constant = 1374
X 10-22 joule per °K.
Absolute temperature °K.
Resistive component of impedance across which thermal noise
is developed.
Frequency band across which
voltage is measured.
In the above equation \f is essentially the
frequency band passed by the intermediate frequency amplifier of the receiver under consideration. This equation can be greatly simplified for the conditions normally encountered
in communications work. If we assume the following conditions: T = 300° K or 27° C or
80.5° F, room temperature; 1f = 8000 cycles
(the average pass band of a communications
receiver or speech amplifier) the equation remicrovolts. Acduces to: Et.m.s. = 0.0115
cordingly, the thermal -agitation voltage appearing in the center of half -wave antenna (assuming effective temperature to be 300° K)
having a radiation resistance of 73 ohms is
151
www.americanradiohistory.com
152
R
-F
Vacuum
Tube
Amplifiers
approximately 0.096 microvolts. Also, the thermal agitation voltage appearing across a 500,000 -ohm grid resistor in the first stage of a
speech amplifier is approximately 8 microvolts
under the conditions cited above. Further, the
voltage due to thermal agitation being impressed on the grid of the first r -f stage in a
receiver by a first tuned circuit whose resonant resistance is 50,000 ohms is approximately
2.5 microvolts. Suffice to say, however, that
the value of thermal agitation voltage appearing across the first tuned circuit when the antenna is properly coupled to this circuit will
be very much less than this value.
It is common practice to match the impedance of the antenna transmission line to the
input impedance of the grid of the first r -f amplifier stage in a receiver. This is the condition of antenna coupling which gives maximum
gain in the receiver. However, when u -h -f tubes
such as acorns and miniatures are used at frequencies somewhat less than their maximum
capabilities, a significant improvement in signal -to -noise ratio can be attained by increasing the coupling between the antenna and first
tuned circuit to a value greater than that which
gives greatest signal amplitude out of the receiver. In other words, in the 10, 6, and 2 meter bands it is possible to attain somewhat improved signal -to -noise ratio by increasing antenna coupling to the point where the gain of
the receiver is slightly reduced.
It is always possible, in addition, to obtain
improved signal -to -noise ratio in a v -h -f receiver through the use of tubes which have
improved input impedance characteristics at
the frequency in question over conventional
types.
The limiting condition for sensitivity in any receiver is the
thermal noise generated in the antenna and in
the first tuned circuit. However, with proper
coupling between the antenna and the grid of
the tube, through the first tuned circuit, the
Noise Factor
noise contribution of the first tuned circuit
can be made quite small. Unfortunately, though,
the major noise contribution in a properly designed receiver is that of the first tube. The
noise contribution due to electron flow and
due to losses in the tube can be lumped into
an equivalent value of resistance which, if
placed in the grid circuit of a perfect tube having the same gain but no noise would give the
same noise voltage output in the plate load.
The equivalent noise resistance of tubes such
as the 6SK7, 6SG7, etc., runs from 5000 to
10,000 ohms. Very high Gm tubes such as the
6AC7 and 6AK5 have equivalent noise resistances as low as 700 to 1500 ohms. The lower
the value of equivalent noise resistance, the
THE
RADIO
lower will be the noise output under a fixed
set of conditions.
The equivalent noise resistance of a tube
must not be confused with the actual input
loading resistance of a tube. For highest signal -to -noise ratio in an amplifier the input
loading resistance should be as high as possible so that the amount of voltage that can be
developed from grid to ground by the antenna
energy will be as high as possible. The equivalent noise resistance should be as low as
possible so that the noise generated by this
resistance will be lower than that attributable
to the antenna and first tuned circuit, and the
losses in the first tuned circuit should be as
low as possible.
The absolute sensitivity of receivers has
been designated in recent years in government
and commercial work by an arbitrary dimensionless number known as "noise factor" or N.
The noise factor is the ratio of noise output
of a "perfect" receiver having a given amount
of gain with a dummy antenna matched to its
input, to the noise output of the receiver under
measurement having the same amount of gain
with the dummy antenna matched to its input.
Although a perfect receiver is not a physically
realizable thing, the noise factor of a receiver
under measurement can be determined by calculation from the amount of additional noise
(from a temperature -limited diode or other calibrated noise generator) required to increase
the noise power output of a receiver by a predetermined amount.
Tube Input
As has been mentioned in a pre -
vious paragraph, greatest gain
in a receiver is obtained when
the antenna is matched, through the r -f coupling transformer, to the input resistance of
the r -f tube. However, the higher the ratio of
tube input resistance to equivalent noise resistance of the tube the higher will be the signal -to -noise ratio of the stage -and of course,
the better will be the noise factor of the overall receiver. The input resistance of a tube
is very high at frequencies in the broadcast
band and gradually decreases as the frequency
increases. Tube input resistance on conventional tube types begins to become an important factor at frequencies of about 25 Mc. and
above. At frequencies above about 100 Mc. the
use of conventional tube types becomes impracticable since the input resistance of the
tube has become so much lower than the equivalent noise resistance that it is impossible
to attain reasonable signal -to -noise ratio on
any but very strong signals. Hence, special
v -h-f tube types such as the 6AK5, 6ÁG5, and
6CB6 must be used.
The lowering of the effective input resistLoading
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HANDBOOK
R
ance of a vacuum tube at higher frequencies
is brought about by a number of factors. The
first, and most obvious, is the fact that the
dielectric loss in the internal insulators, and
in the base and press of the tube increases
with frequency. The second factor is due to
the fact that a finite time is required for an
electron to move from the space charge in the
vicinity of the cathode, pass between the grid
wires, and travel on to the plate. The fact that
the electrostatic effect of the grid on the moving electron acts over an appreciable portion
of a cycle at these high frequencies causes a
current flow in the grid circuit which appears
to the input circuit feeding the grid as a resistance. The decrease in input resistance of
a tube due to electron transit time varies as
the square of the frequency. The undesirable
effects of transit time can be reduced in certain cases by the use of higher plate voltages.
Transit time varies inversely as the square
root of the applied plate voltage.
Cathode lead inductance is an additional
cause of reduced input resistance at high frequencies. This effect has been reduced in certain tubes such as the 6S117 and the 6AK5 by
providing two cathode leads on the tube base.
One cathode lead should be connected to the
input circuit of the tube and the other lead
should be connected to the by -pass capacitor
for the plate return of the tube.
The reader is referred to the Radiation Laboratory Series, Volume 23: "Microwave Receivers" (McGraw -Hill, publishers) for additional
information on noise factor and input loading
of vacuum tubes.
Amplifiers
-F
153
OA
AMPLIFICATION AT RESONANCE (APPROX.) =GMWLQ
OB
AMPLIFICATION AT RESONANCE (APPROX ) =GWMQ
© AMPLIFICATION
AT RESONANCE(APPRO[kGMK
U)
K2t-1-s
1
QP
WHERE
S
PRI. ANO SEC. RESONANT AT SAME FREQUENCY
2 K IS COEFFICIENT OF COUPLING
1.
IF FRI. AND SEC. Q ARE APPROXIMATELY THE SAME.
8 -2
TOTAL BANDWIDTH
1.2 K
CENTER FREQUENCY
MAXIMUM AMPLITUDE OCCURS AT CRITICAL COUPLING
Plate- Circuit
Considerations
WHEN
-
KQP
Noise is generated in a vacuum tube by the
fact that the current flow within the tube is not
a smooth flow but rather is made up of the continuous arrival of particles (electrons) at a
very high rate. This shot effect is a source of
noise in the tube, but its effect is referred
back to the grid circuit of the tube since it is
included in the equivalent noise resistance
discussed in the preceding paragraphs.
For the purpose of this section,
it will be considered that the
function of the plate load circuit of a tuned vacuum -tube amplifier is to deliver energy to the next stage with the greatest
Plate Circuit
Coupling
efficiency over the required band of frequencies. Figure 1 shows three methods of inter stage coupling for tuned r -f voltage amplifiers.
In figure IA omega (w) is 2n times the resonant frequency of the circuit in the plate of
Figure
Gain
1
equations for pentode r -f amplifier
stages operating into a tuned load
the amplifier tube, and L and Q are the inductance and Q of the inductor L. In figure 1B the
notation is the same and M is the mutual inductance between the primary coil and the secondary coil. In figure 1C the notation is again
the same and k is the coefficient of coupling
between the two tuned circuits. As the coefficient of coupling between the circuits is
increased the bandwidth becomes greater but
the response over the band becomes progressively more double -humped. The response over
the band is the most flat when the Q's of primary and secondary are approximately the same
and the value of each Q is equal to 1.75/k.
www.americanradiohistory.com
154
R
-F
Variable -Mu Tubes
Vacuum
Tube
Amplifiers
It is common practice to
control the gain of a succession of r -f or i -f amplifier stages by varying the average bias on
their control grids. H)wever, as the bias is
raised above the operating value on a conventional sharp- cutoff tube the tube becomes increasingly non -linear in operation as cutoff of
plate current is approached. The effect of such
non -linearity is to cause cross modulation between strong signals which appear on the grid
of the tube. When a tube operating in such a
manner is in one of the first stages of a receiver a number of signals are appearing on its
grid simultaneously and cross modulation between them will take place. The result of this
effect is to produce a large number of spurious
signals in the output of the receiver -in most
in
R
-F Stages
THE
RADIO
cases these signals will carry the modulation
of both the carriers which have been cross
modulated to produce the spurious signal.
The undesirable effect of cross modulation
can be eliminated in most cases and greatly
reduced in the balance through the use of a
variable -mu tube in all stages which have a -v-c
voltage or other large negative bias applied to
their grids. The variable -mu tube has a characteristic which causes the cutoff of plate current to be gradual with an increase in grid
bias, and the reduction in plate current is accompanied by a decrease in the effective amplification factor of the tube. Variable -mu tubes
ordinarily have somewhat reduced Gm as compared to a sharp- cutoff tube of the same group.
Hence the sharp- cutoff tube will perform best
in stages to which a-v -c voltage is not applied.
RADIO- FREQUENCY POWER AMPLIFIERS
All modern transmitters in the medium -frequency range and an increasing percentage of
those in the v -h -f and u -h-f ranges consist of
a comparatively low -level source of radio-frequency energy which is multiplied in frequency
and successively amplified to the desired power
level. Microwave transmitters are still predominately of the self- excited oscillator type, but
when it is possible to use r -f amplifiers in
s -h -f transmitters the flexibility of their application will be increased. The following portion of this chapter will be devoted, however,
to the method of operation and calculation of
operating characteristics of r-f power amplifiers for operation in the range of approximately 3.5 to 500 Mc.
8 -3
Class C R -F
Power Amplifiers
The majority of r -f power amplifiers fall into
the Class C category since such stages can
be made to give the best plate circuit efficiency of any present type of vacuum-tube amplifier. Hence, the cost of tubes for such a stage
and the cost of the power to supply that stage
is least for any given power output. Nevertheless, the Class C amplifier gives less power
gain than either a Class A or Class B amplifier under similar conditions since the grid of
a Class C stage must be driven highly positive over the portion of the cycle of the exciting wave when the plate voltage on the amplifier is low, and must be at a large negative
potential over a large portion of the cycle so
that no plate current will flow except when
plate voltage is very low. This, in fact, is the
fundamental reason why the plate circuit efficiency of a Class C amplifier stage can be
made high -plate current is cut off at all times
except when the plate -to- cathode voltage drop
across the tube is at its lowest value. Class
C amplifiers almost invariably operate into a
tuned tank circuit as a load, and as a result
are used as amplifiers of a single frequency
or of a comparatively narrow band of frequencies.
2 shows the relation ships between the various
voltages and currents over
one cycle of the exciting grid voltage for a
Class C amplifier stage. The notation given in
figure 2 and in the discussion to follow is the
same as given at the first of Chapter Six under "Symbols for Vacuum -Tube Parameters."
The various manufacturers of vacuum tubes
publish booklets listing in adequate detail alternative Class C operating conditions for the
tubes which they manufacture. In addition,
operating condition sheets for any particular
type of vacuum tube are available for the asking from the different vacuum -tube manufacturers. It is, nevertheless, often desirable to
determine optimum operating conditions for a
tube under a particular set of circumstances.
To assist in such calculations the following
paragraphs are devoted to a method of calculating Class C operating conditions which is
moderately simple and yet sufficiently accurate for all practical purposes.
Relationships in
Class
C
Stage
www.americanradiohistory.com
Figure
Class
HANDBOOK
R
C
-F Amplifiers
155
tional grid voltage -plate current operating
curves, the calculation is considerably simplified if the alternative "constant- current
curve" of the tube in question is used. This is
true since the operating line of a Class C amplifier is a straight line on a set of constantcurrent curves. A set of constant -current curves
on the 250TH tube with a sample load line
drawn thereon is shown in figure 5.
In calculating and predicting the operation
PLATE
VOLTAGE
EPM
EBB
P
eP
O1._.L.L- --'-- -1-I
--- r
I
1
PEAK
PLATE
CURRENT
_
I
If--t- - -
--I--
-I
I19P-.1.-9P+11
I
I
I
I
I
I
7I
I
I
t
I
11
VI
1
I
r
I
I
EGM
e
-
I
I
II
II
I
I
I
I
-I
I
-
V
t1
III
-
I
III
11
III
0i- d11i
Ecc
-i
FUNDAMENTAL COMPONENT
OF PLATE CURRENT
'I
I
IG MAS.
1
:
I
{
-1-,,,*
I
I
-
I
I
I
1------ I-
I11
I
III
111
I
I
III
GRID
III VOLTAGE
11-- -I
G
'-Ieca-1--
Figure
Instantaneous electrode
voltages and currents for
I
2
and
a
amplifier
Calculation of Class
C Amplifier Operating
Characteristics
GRID
I
FI-CURRENT
tank
Class
circuit
C r -f
power
Although Class C opcrating conditions can
be determined with the
aid of the more conven-
of a vacuum tube as a Class C radio -frequency
amplifier, the considerations which determine
the operating conditions are plate efficiency,
power output required, maximum allowable
plate and grid dissipation, maximum allowable
plate voltage and maximum allowable plate
current. The values chosen for these factors
will depend both upon the demands of a par-
ticular application and upon the tube chosen.
The plate and grid currents of a Class C
amplifier tube are periodic pulses, the durations of which are always less than 180 degrees. For this reason the average grid current, average plate current, power output, driving power, etc., cannot be directly calculated
but must be determined by a Fourier analysis
from points selected at proper intervals along
the line of operation as plotted upon the constant- current characteristics. This may be done
either analytically or graphically. While the
Fourier analysis has the advantage of accuracy, it also has the disadvantage of being
tedious and involved.
The approximate analysis which follows
has proved to be sufficiently accurate for most
applications. This type of analysis also has
the advantage of giving the desired information at the first trial. The system is direct in
giving the desired information since the important factors, power output, plate efficiency,
and plate voltage are arbitrarily selected at
the beginning.
first step in the method to
described is to determine the
power which must be delivered
by the Class C amplifier. In making this determination it is well to remember that ordinarily
from 5 to 10 per cent of the power delivered
by the amplifier tube or tubes will be lost in
well- designed tank and coupling circuits at
frequencies below 20 Mc. Above 20 Mc. the
tank and circuit losses are ordinarily somewhat above 10 per cent.
The plate power input necessary to produce
the desired output is determined by the plate
efficiency: Pin = Pout/Np.
For most applications tt is desirable to operate at the highest practicable efficiency. High efficiency operation usually requires less expensive tubes and power supplies, and the
Method of
The
Calculation
be
www.americanradiohistory.com
156
R
-F
Vacuum
Tube
\
Amplifiers
THE
i'\ E
\\
agi
E \EE
EE
\u
=\
MENE
7.0
E
.0
E
\1
O
7.0
E
EEI \
.0
40
30
RADIO
tt
e
RATIO
,.
iáu
t
3.0
I
-10
EEE IMEN
-20
-10
.1
RATIO
Figure 3
Relationship between the peak value of the
fundamental component of the tube plate current, and average plate current; as compared
to the ratio of the instantaneous peak value
of tube plate current, and average plate
current
amount of artificial cooling required is frequently less than for low- efficiency operation.
On the other hand, high- efficiency operation
usually requires more driving power and involves the use of higher plate voltages and
higher peak tube voltages. The better types
of triodes will ordinarily operate at a plate
efficiency of 75 to 85 per cent at the highest
rated plate voltage, and at a plate efficiency
of 65 to 75 per cent at intermediate values of
plate voltage.
The first determining factor in selecting a
tube or tubes for a particular application is
the amount of plate dissipation which will be
required of the stage. The total plate dissipation rating for the tube or tubes to be used in
the stage must be equal to or greater than that
calculated from: Pp = Pin - Pout.
After selecting a tube or tubes to meet the
power output and plate dissipation requirements it becomes necessary to determine from
the tube characteristics whether the tube selected is capable of the desired operation and,
if so, to determine the driving power, grid
bias, and grid dissipation.
The complete procedure necessary to determine a set of Class C amplifier operating conditions is given in the following steps:
1. Select the plate voltage, power output,
and efficiency.
Figure
4
Relationship between the ratio of the peak
value of the fundamental component of the
grid excitation voltage, and the overage grid
bias; as compared to the ratio between instantaneous peak grid current and average
grid current
2.
Determine plate input from: Pin
=
Pout/Np.
3 Determine plate dissipation
Pp= Pin - Pout Pp must
from:
not exceed
maximum rated plate dissipation for tube
or tubes selected.
4. Determine average plate current from:
lb = Pin /Ebb
5.
Determine approximate
;p.a.
from:
4.9 lb for Np = 0.85
tpmas 4.5 lb for Np = 0.80
tpmas = 4.0 'b for N = 0.75
tpmax= 3.51b for Np =0.70
tpmax
6. Locate
=
=
the point on
constant -current
characteristics where the constant plate
current line corresponding to the approximate ipmax determined in step 5
crosses the line of equal plate and grid
voltages (diode line). Read epmin at this
point. In a few cases the lines of constant plate current will inflect sharply
upward before reaching the diode line.
In these cases epmin should not be read
at the diode line but at the point where
the plate current line intersects a line
drawn from the origin through these
points of inflection.
www.americanradiohistory.com
FIRST TRIAL POINT
-s,
N
Om
157
Constant Current Calculations
HANDBOOK
FINAL POINT
EIMAC 250TH
CONSTANT CURRENT
CHARACTERISTICS
....
sa:
pP:
.
,
r-;pze
o
ó
-_-.
00
_
Ti
ERE
7b.
....... .........
,
MOO
EGO=
- 240
LOAD LINE
x00
XOD
Ebb =4-3500
PLATE VOLTAGE -VOLTS
FIGURE
5
Active portion of the operating load line for an Eimoc 250TH Class C r -f power amplifier,
showing first trial point and the final operating point
7. Calculate Epm from: Epm = Ebb
- epmin
13.
8. Calculate the ratio Ipm /lb from:
1pm
lb
2
Ecc
1
Epm
Fos
Calculate a new value for ipmax from
the ratio found in step 9.
tpm as = (ratio from step 9) lb
Ecc
f3p
=2.32(
Ipm
Ib
-
-
µ
egmp)
Ebb
fi
1
X
L
for tetrodes, where
camp cos
O
-
En,
J
1112
tt
is the grid- screen
amplification factor, and Ec2 is the d -c
screen voltage.
and 10.
cos
\
1- cos 6p
constant current characteristics for the values of
epmin and ipmax determined in steps 6
Calculate the cosine of one -half the
angle of plate current flow from:
Epm
Op
X
- cos Op
for triodes.
11. Read egmp and igmax from the
12.
1
-
Np Ebb
9. From the ratio of Ipm /Ib calculated in
step 8 determine the ratio ipmax /Ib from
figure 3.
10.
Calculate the grid bias voltage from:
14.
Calculate the peak fundamental grid excitation voltage from:
Earn = egmp
-
Ecc
1.57)
15.
Calculate the ratio Egm /Ecc for the val-
www.americanradiohistory.com
158
R
-F
Vacuum
Amplifiers
Tube
ues of Ecc and Egm found in steps 13
12.
and 14.
Bp=2.32(1.73-1.57)=0.37
ratio
13.
1
-
- 0.37
X
(_ 3240
[0.37
3500
240)
37
37
14.
- 240 volts
Egm = 240 - ( -240) = 480 volts grid
15.
Egm /Ecc =
16.
igmax /Ic = 5.75 (from figure 4)
17.
le = 0.430/5.75 = 0.075 amp. (75 ma.
grid current)
18.
Pd = 0.9X480X0.075 = 32.5 watts
driving power
=
tgmax
Ratio from step
Ecc
1
Calculate the average grid current from
the ratio found in step 16, and the value
of igmax found in step 11:
IC
cos
RADIO
(9p = 68.3°)
16. Read igmax /Ic from figure 4 for the
Egm /Ecc found in step 15.
17.
THE
16
swing
18. Calculate approximate grid driving pow-
er from:
Pd
0.9 Egmlc
=
Calculate grid dissipation from:
Pa = Pa + Eccic
19.
Pg must not exceed the maximum rated
grid dissipation for the tube selected.
Sample
typical example of a Class C
amplifier calculation is shown
A
in the example below. Reference
3, 4 and 5 in the calcula-
is made to figures
tion.
1.
Desired power output -800 watts.
2.
Desired plate voltage -3500 volts.
Desired plate efficiency -80 per cent
(Np = 0.80)
Pin = 800 /0.80 = 1000 watts
3.
The power output of any type of r -f ampli-
fier is equal to:
Ib
-
It is frequently of importance to know the
value of load impedance into which a Class
C amplifier operating under a certain set of
conditions should operate. This is simply R L_
Epm /Ipm. In the case of the operating conditions just determined for a 250TH amplifier
stage the value of load impedance is:
RL _
Approximate ipmax = 0.285 X 4.5
= 1.28 ampere
trial point)
7.
Epm = 3500
volts (see figure
- 260 = 3240
5
11.
egmp = 240 volts
igmax = 0.430 amperes
3240
Ipm
.495
6600 ohms
-Xlb
lb
of Amplifier
Tank Circuit
volts
(Both above from final point on figure
Epm
Ipm =
In order to obtain good plate
tank circuit tuning and low
radiation of harmonics from
an amplifier it is necessary that the plate tank
circuit have the correct Q. Charts giving compromise values of Q for Class C amplifiers
are given in the chapter, Generation o/ R -F
Energy. However, the amount of inductance
Q
9. ipmax /Ib = 4.1 (from figure 3)
ipmax = 0.285X4.1 = 1.17
-- -Ipm
first
8. Ipm /lb = 2X0.80X3500/3240 =
5600/3240 = 1.73
10.
-
ratio determined in step 8 above (in this type
of calculation) by multiplying this ratio times
Pp = 1000
800 = 200 watts
Use 250TH; max. Pp = 250w;µ = 37.
6. epmin = 260
2
IpmEpm /2 = Po
Ipm can be determined, of course, from the
4. lb = 1000/3500 = 0.285 ampere (285 ma.)
Max. Ib for 250TH is 350 ma.
5.
- 240 = -
(- 240X0.75) = 14.5 watts
grid dissipation
Max. Pg for 250TH is 40 watts
19. Pg = 32.5
Calculation
480/
5)
required for a specified tank circuit Q under
specified operating conditions can be calculated from the following expression:
www.americanradiohistory.com
Class
HANDBOOK
R
B
Q
Quick Method of
Calculating Amplifier
Plate Efficiency
The plate circuit efficiency of a Class B or
Class C r -f amplifier
can be determined from
the following facts. The plate circuit efficiency
of such an amplifier is equal to the product of
two factors,
which is equal to the ratio of
Epm to Ebb (F, = Epm /Ebb) and
which is
proportional to the one -half angle of plate current flow, 0p. A graph of F, against both 0,
and cos Bp is given in figure 6. Either 0p or
cos Bp may be used to determine F,. Cos 0p
may be determined either from the procedure
previously given for making Class C amplifier
computations or it may be determined from the
following expression:
F
F
Bp =
-
Ecc
+
Ebb
µEBm-Epm
Example of
Method
is desired to know the one -half
angle of plate current flow and
It
the plate circuit efficiency for
an 812 tube operating under the following conditions which have been assumed from inspection of the data and curves given in the RCA
Transmitting Tube Handbook HB -3:
1.
Ebb = 1100 volts
Ecc = -40 volts
1.'
....,....
.......,....
.......-....
...\\...
Ism
= 2 ir X operating frequency
= Tank inductance
= Required tube load impedance
= Effective tank circuit Q
tank circuit Q of 12 to 20 is recommended
for all normal conditions. However, if a balanced push -pull amplifier is employed the tank
receives two impulses per cycle and the circuit Q may be lowered somewhat from the
above values.
cos
DN,IIII.`....MM....
MN
A
159
R\......
Ri,
L
RL
Q
Amplifiers
-F
aee1
F2
aee
o.
i.........,,..
I..........,.
awl
D
ore
ore
ar211..........,
.... .....
aM
0
,
10
20
30
PP
IN
40
50
e0
70
e0
90
100 110
120
ELECTRICAL DEGREES
. .». <.. .... .
.
.:,. ... <.,. >... -..,»
I
cos
I
AP
Figure 6
Relationship between Factor F_ and the
half -angle of plate current flow in an amplifier with sine -wave input and output voltage,
operating at a grid -bias voltage greater than
cut -off
5. Np= F, X F, = 0.91 X 0.79 = 0.72
(72 per cent efficiency)
F, could be called the plate -voltage-swing
efficiency factor, and F2 can be called the
operating -angle efficiency factor or the maximum possible efficiency of any stage running
with that value of half -angle of plate current
flow.
Np is, of course, only the ratio between
power output and power input. If it is desired
to determine the power input, exciting power,
and grid current of the stage, these can be obtained through the use of steps 7, 8, 9, and 10
of the previously given method for power inis
put and output; and knowing that
0.095 ampere the grid circuit conditions can
be determined through the use of steps 15, 16,
17, 18 and 19.
= 29
Ea.
= 120 volts
Epm = 1000 volts
2.
F,
3.
cos 0p=
=
Epm/Ebb
=
8 -4
0.91
- 29 X 40 + 1100
29 X 120 - 1000
60
- 0.025
2480
4.
F2= 0.79 (by reference to figure 6)
Class B Radio
Frequency Power Amplifiers
Radio frequency power amplifiers operating
under Class B conditions of grid bias and excitation voltage are used in two general types
of applications in transmitters. The first general application is as a buffer amplifier stage
where it is desired to obtain a high value of
power amplification in a particular stage. A
particular tube type operated with a given
plate voltage will be capable of somewhat
greater output for a certain amount of excitation power when operated as a Class B ampli-
www.americanradiohistory.com
160
R
-F
Tube
Vacuum
Amplifiers
Ecz= 7-400
l
RADIO
THE
V.
Ec3=0v.
Eci=
II
Ec=+Go
__A
___
rd.
1
}s0
Eci=+4o
i
I
\t
Eci= +20
Eci=
I
1
i
I
I
iii
/
c2,Ec=+ioo
J--
Icz Ec=+w
l00
200
300
400
I
Fri,
_,--.-500
G00
700
-20
Eci=-4o
G0o
am
000
um,
PLATE VOLTS
.Hr,
v.
A.....
._....
..,.,.
...
Figure 7
AVERAGE PLATE CHARACTERISTICS OF 813 TUBE
fier than when operated as
fier.
a
Class C ampli-
Calculation of
Calculation of the operating
conditions for this type of
Characteristics
Class B r -f amplifier can be
carried out in a manner similar to that described in the previous paragraphs, except that the grid bias voltage is set
on the tube before calculation at the value:
Ecc =
Ebb /IL. Since the grid bias is set at
cutoff the one-half angle of plate current flow
is 900; hence cos
is fixed at 0.00. The
plate circuit efficiency for a Class B r -f amplifier operated in this manner can be determined in the following manner:
Operating
NP
=78.5
(
-1
Epm
"Class
Linear"
B
plication.
Calculation of Operoting Parameters for a
Class B Linear Amplifier
Ebb //
The
calculated. Then, with the exciting voltage
reduced to one -half for the no- modulation condition of the exciting wave, and with the same
value of load resistance reflected on the tube,
the plate input and plate efficiency will drop
to approximately one-half the values at the
100 per cent positive modulation peak and the
power output of the stage will drop to onefourth the peak- modulation value. On the negative modulation peak the input, efficiency, and
output all drop to zero.
In general, the proper plate voltage, bias
voltage, load resistance and power output
listed in the tube tables for Class B audio
work will also apply to Class B linear r -f ap-
The second type of Class B
r-f amplifier is the so- called
Class Il linear amplifier which
is often used in transmitters for the amplification of a single - sideband signal or a conventional amplitude- modulated wave. Calculation
of operating conditions may be carried out in
a manner similar to that previously described
with the following exceptions: The first trial
operating point is chosen on the basis of the
100 per cent positive modulation peak of the
modulated exciting wave. The plate circuit
and grid peak voltages and currents can then
be determined and the power input and output
7 illustrates
characteristic
curves for an 813
Figure
the
tube.
Assume
the
plate supply to be 2000 volts, and the screen
supply to be 400 volts. To determine the operating parameters of this tube as a Class B linear r -f amplifier, the following steps should
be taken:
1.
The grid bias is chosen so that the resting plate current will produce approximately 1/3 of the maximum plate dissipation of the tube. The maximum dissipation of the 813 is 125 watts, so the
bias is set to allow one -third of this
value, or 42 watts of resting dissipation.
At a plate potential of 2000 volts, a
www.americanradiohistory.com
HANDBOOK
2.
Linear Amplifier Parameters
plate current of 21 milliamperes will
produce this figure. Referring to figure
7, a grid bias of -45 volts is approximately correct.
A practical Class 13 linear r -f amplifier
runs at an efficiency of about 66% at full
output, the efficiency dropping to about
33% with an unmodulated exciting signal. In the case of single- sideband suppressed carrier excitation, a no- excitation condition is substituted for the unmodulated excitation case, and the linear amplifier runs at the resting or quiescent input of 42 watts with no exciting
signal. The peak allowable power input
to the 813 is:
Input Peak Power (Wp) _
(watts)
Plate Dissipation X 100
(100
-
BO
`
GO
Ec3=ov.
Eci=+ioov.
ci=rsov.,
40
¡ Ec-raov.
Rib_ Eu=+sov
xo
Ecr+20
o
-
_
-
.
200
100
Eg
VS.
E
x
P
Ep
-
400
_
x
-= 0.189 ampere
2000
0.189
1580
=
=
0.5 x .189
6000 ohms
9. If a loaded plate tank circuit Q of 12 is
desired, the reactance of the plate tank
capacitor at the r e s on an t frequency
The plate current flow of the linear amplifier is 1800, and the plate current
pulses have a peak of 3.14 times the
maximum signal current:
3.14
resistance is:
epmin
379
Wp
Ep
-
0.5ipmax
The maximum signal plate current is:
tpmax =
should be:
Reactance (ohms) =
0.595 ampere
RL
--
Referring to figure 7, a current of 0.605
ampere (Point A) will flow at a positive
grid potential of 60 volts and a minimum
plate potential of 420 volts. The grid is
biased at -45 volts, so a peak r -f grid
voltage of 60+45 volts = 105 volts is required.
The grid driving power required for the
B linear stage may be found by the
aid of figure 8. It is one -quarter the product of the peak grid current times the
peak grid voltage:
Class
0.02
X 105
Pp =
10.
500 ohms
For an operating frequency of 4.0 Mc.,
the effective resonant capacity is:
106
C=
=
6.28
11.
x
4.0
x
80 µµtd.
500
The inductance required to resonate at
4.0 Mc. with this value of capacity is:
500
L
- 0.53
6000
=
12
Q
6.
300
Figure 8
CHARACTERISTICS OF 813
TUBE
8. The plate load
100 = 379 watts
RL
5.
E
PLATE VOLTS
33
4.
ECO +400 V.
-% plate efficiency)
125
3.
161
watt
6.28
=
x
4.0
19.9 microhenries
4
7.
The single tone power output of the 813
stage is:
Pp = 78.5 (Ep - epmin) x Ip
Pp = 78.5 (2000 - 420) x .189 = 235 watts
Grid Circuit
The maximum positive grid
potential is 60 volts, and
the peak r -f grid voltage is
105 volts. Required driving power is 0.53 watt.
The equivalent grid resistance of this stage is:
Considerations
www.americanradiohistory.com
1.
162
R
-F
Tube
Vacuum
Rig
(e5)2
-
1052
-
2XPg 2X0.53
Amplifiers
As in the case of the Class B audio amplifier the grid resistance of the linear
amplifier varies from infinity to a low
value when maximum grid current is
drawn. To decrease the effect of this
resistance excursion, a swamping resistor should be placed across the grid tank
circuit. The value of the resistor should
be dropped until a shortage of driving
power begins to be noticed. For this example, a resistor of 3,000 ohms is used.
The grid circuit load for no grid current
is now 3,000 ohms instead of infinity,
and drops to 2400 ohms when maximum
grid current is drawn.
3.
is chosen for the grid
tank. The capacitive reactance required
A
circuit
of
Q
15
is:
-=
2400
X
=
160 ohms
15
4. At 4.0 Mc. the effective capacity is:
106
=
C=
248 µµEd.
6.28x4X154
5. The inductive reactance required to reso-
nate the grid circuit at 4.0 Mc. is:
160
L=
=
6.4 microhenries
6.28 x 4.0
6.
substituting the loaded grid resistance figure in the formula in the first
paragraph, the grid driving power is now
found to be approximately 2.3 watts.
By
Screen Circuit
Considerations
Special
8 -5
-
10,400 ohms
2.
THE
reference to the plate
characteristic curve of the
By
813 tube, it can be seen that
at a minimum plate potential of 500 volts, and
a maximum plate current of 0.6 ampere, the
screen current will be approximately 30 milliamperes, dropping to one or two milliamperes
in the quiescent state. It is necessary to use
a well -regulated screen supply to hold the
screen voltage at the correct potential over
this range of current excursion. The use of an
electronic regulated screen supply is recommended.
R
RADIO
-F Power
Amplifier Circuits
The r-f power amplifier discussions of Sections 8 -4 and 8 -5 have been based on the assumption that a conventional grounded- cathode
or cathode -return type of amplifier was in question. It is possible, however, as in the case of
a -f and low-level r -f amplifiers to use circuits
in which electrodes other than the cathode are
returned to ground insofar as the signal potential is concerned. Both the plate-return or
cathode -follower amplifier and the grid- return
or grounded -grid amplifier are effective in certain circuit applications as tuned r -f power
amplifiers.
Disadvantages of
Grounded -Cothode
Amplifiers
An
undesirable aspect of
the operation of cathode return r -f power amplifiers
using triode tubes is that
such amplifiers must be neutralized. Principles and methods of neutralizing r -f power amplifiers are discussed in the chapter Generation of R -F Energy. As the frequency of operation of an amplifier is increased the stage becomes more and more difficult to neutralize
due to inductance in the grid and plate leads
of the tubes and in the leads to the neutralizing capacitors. In other words the bandwidth
of neutralization decreases as the frequency
is increased. In addition the very presence of
the neutralizing capacitors adds additional
undesirable capacitive loading to the grid and
plate tank circuits of the tube or tubes. To
look at the problem in another way, an amplifier that may be perfectly neutralized at a frequency of 30 Mc. may be completely out of
neutralization at a frequency of 120 Mc. Therefore, if there are circuits in both the grid and
plate circuits which offer appreciable impedance at this high frequency it is quite possible that the stage may develop a "parasitic
oscillation" in the vicinity of 120 Mc.
This condition of restricted range neutralization of r -f
power amplifiers can be greatly alleviated through the use of a cathode return or grounded -grid r -f stage. The grounded grid amplifier has the following advantages:
Grounded -Grid
R-F Amplifiers
1.
The output capacitance of a stage is reduced to approximately one -half the value
which would be obtained if the same tube
or tubes were operated as a conventional
neutralized amplifier.
2.
The tendency toward parasitic oscillations
in such a stage is greatly reduced since
the shielding effect of the control grid be-
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HANDBOOK
Amplifier
Grounded Grid
tween the filament and the plate is effective over a broad range of frequencies.
The feedback capacitance within the stage
3.
is the plate -to- cathode capacitance which
is ordinarily very much less than the gridto -plate capacitance. Hence neutralization
is ordinarily not required. If neutralization
is required the neutralizing capacitors are
very small in value and are cross connected between plates and cathodes in a
push -pull stage, or between the opposite
end of a split plate tank and the cathode
in a single -ended stage.
The disadvantages of a grounded -grid amplifier are:
1. A large amount of excitation energy is required. However, only the normal amount
of energy is lost in the grid circuit of the
amplifier tube; all additional energy over
this amount is delivered to the load circuit as useful output.
2. The cathode of a grounded -grid amplifier
stage is "hot" to r.f. This means that the
cathode must be fed through a suitable impedance from the filament supply, or the
secondary of the filament transformer must
be of the low- capacitance type and adequately insulated for the r -f voltage which
will be present.
3. A grounded -grid r -f amplifier cannot be
plate modulated 100 per cent unless the
output of the exciting stage is modulated
also. Approximately 70 per cent modulation of the exciter stage as the final stage
is being modulated 100 per cent is recommended. However, the grounded -grid r -f
amplifier is quite satisfactory as a Class
B linear r -f amplifier for single sideband
or conventional amplitude modulated waves
or as an amplifier for a straight c -w or
FM signal.
Figure 9 shows a simplified representation
of a grounded -grid triode r -f power amplifier
stage. The relationships between input and
out put power and the peak fundamental components of electrode voltages and currents are
given below the drawing. The calculation of
the complete operating conditions for a
grounded-grid amplifier stage is somewhat more
complex than that for a conventional amplifier
because the input circuit of the tube is in
series with the output circuit as far as the
load is concerned. The primary result of this
effect is, as stated before, that considerably
more power is required from the driver stage.
The normal power gain for a g -g stage is from
3 to 15 depending upon the grid circuit conditions chosen for the output stage. The higher
the grid bias and grid swing required on the
163
c-
{E4NEnr) IPr
PONEN OUTPUT TO LOAD
POWER DEL,ORSED
n
POWER PROM DO UER TO LOAD
TOTAL PONE, DELIVERED BY
,SORKD
DIVE,
EON
5Y OUTPUT TUNE
(IPI+ I4r)
E
MD NO
.!S
E4r
Z.
(PP,o..N
E4r IPr
i
E4 =IPr
o,
POOR,
EPr IP r
Irr
Err
OUTPUT TUBE
Irr
Or E4r lc
SUPPLY
I4r
O,OrE4NIt
E4r
I.N
I
lc
Figure
9
GROUNDED -GRID CLASS
B
OR
CLASS
C
AMPLIFIER
The equations in the above figure give the
relationships between the fundamental com-
ponents of grid and plate potential and current, and the power input and power output
of the stage. An expression for the approximate cathode impedance is given
output stage, the higher will be the requirement from the driver.
Calculation of Operating
Conditions of Grounded
Grid R -F Amplifiers
It is most convenient
to determine the op-
erating conditions for
a Class B or Class C
grounded -grid r -f power amplifier in a two -step
process. The first step is to determine the
plate- circuit and grid- circuit operating conditions of the tube as though it were to operate
as a conventional cathode- return amplifier
stage. The second step is then to add in the
additional conditions imposed upon the operating conditions by the fact that the stage is
to operate as a grounded -grid amplifier.
For the first step in the calculation the procedure given in Section 8 -3 is quite satisfactory and will be used in the example to follow.
Suppose we take for our example the case of a
type 304TL tube operating at 2700 plate volts
at a kilowatt input. Following through the procedure previously given:
1.
Desired power output -850 watts
Desired Plate voltage -2700 volts
Desired plate efficiency -85 per cent
(Np = 0.85)
www.americanradiohistory.com
164
2.
3.
4.
-F
R
Tube Amplifiers
Vacuum
P1° = 850/0.85
=
F,
1000 watts
850 = 150 watts
= 1000
Type 304TL chosen; max. P,
watts, /I= 12.
= 300
Ib = 1000/2700 = 0.370 ampere
(370 ma.)
5.
Approximate ipma, = 4.9 X 0.370 = 1.81
ampere
6.
epm;n= 140 volts (from 30411. con-
stant- current curves)
7.
Epm = 2700 - 140 = 2560 volts
8.
1./lb
9.
0.85 X 2700/2560 = 1.79
= 2 X
igmax/Ib
= 4.65 (from
igmax = 4.65 X 0.370 = 1.72 amperes
11.
egmp = 140 volts
12.
Cos
Op =
=
Op
= Epm /Ebb =
2560/2700 = 0.95
of 59° (from figure 6) = 0.90
Np = F, X F2 = 0.95 X 0.90 = Approx.
0.85 (85 per cent plate efficiency)
Now, to determine the operating conditions
as a grounded -grid amplifier we must also know
the peak value of the fundamental components
of plate current. This is simply equal to
(Ipm /Ib) lb, or:
Ipm = 1.79 X 0.370 = 0.660 amperes (from 4
and 8 above)
The total average power required of the
driver (from figure 9) is equal to Egmlpm /2
(since the grid is grounded and the grid swing
appears also as cathode swing) plus Pd which
is 27.5 watts from 18 above. The total is:
0.480 amperes
= 2.32
(1.79 -1.57)
=
525 X 0.660
Total drive
=
-
172.5 watts
2
figure 3)
10.
igma,
RADIO
F2 for Op
-
P,
THE
0.51
59°
plus 27.5 watts or 200 watts
Therefore the total power output of the stage
is equal to 850 watts (contributed by the
304TL) plus 172.5 watts (contributed by the
driver) or 1022.5 watts. The cathode driving
impedance of the 30411. (again referring to
figure 7) is approximately:
Zk = 525/(0.660 + 0.116) = approximately 675
ohms.
1
13.
Ecc-
-
X
1- 0.51
[0.51 (
`
_
2560
-
140)
12
-
Plate- Return or
Cathode- Follower
2700
12
J
-385 volts
14.
Egm = 140 -( -385) = 525
15.
Egm /Ecc =
16.
igmax /Ia = approx. 8.25 (extrapolated
volts
-1.36
from figure 4)
17.
la = 0.480/8.25
grid current)
18.
Pd = 0.9
19.
Pg
=
Max.
X 525 X
27.5
P,
=
0.058 (58 ma. d -c
0.058 = 27.5 watts
-( -385 X 0.058) = 5.2
watts
for 304TL is 50 watts
We can check the operating plate efficiency
of the stage by the method described in Section 8 -4 as follows:
Power Amplifier
Circuit
R
-F
diagram,
elec-
trodepotentials and currents, an d operating
conditions for a cathode- follower r -f power amplifier are given in
figure 10. This circuit can be used, in addition to the grounded -grid circuit just discussed, as an r -f amplifier with a triode tube
and no additional neutralization circuit. However, the circuit will oscillate if the impedance from cathode to ground is allowed to become capacitive rather than inductive or resistive with respect to the operating frequency. The circuit is not recommended except for
v -h -f or u -h -f work with coaxial lines as tuned
circuits since the peak grid swing required
on the r -f amplifier stage is approximately
equal to the plate voltage on the amplifier
tube if high- efficiency operation is desired.
This means, of course, that the grid tank must
be able to withstand slightly more peak voltage than the plate tank. Such a stage may not
be plate modulated unless the driver stage is
modulated the same percentage as the final
amplifier. However, such a stage may be used
as an amplifier or modulated waves (Class B
linear) or as a c -w or FM amplifier.
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HANDBOOK
POWER OUTPUT TO LOAD
G
EpM
(
POWER FROM DRIVER TO LOAD
TOTAL POWER FROM DRIVER_
T.
PM. IGM)
EPM IPM
2
EPM IOM
2
(EPM +eGMP) IGM
ECU IGM
z
2
(EPM + eGMP)
AppROA
ASSUMING IGM
y
I
e
IC
c APPROA
ZG
APPROA.
I GM
O S
(Ecc
eoup) lc
( EPM+ eGMP
I
I
Figure 10
CATHODE -FOLLOWER
R
-F
Control Grid Dissipation
in Grounded -Grid Stages
Tetrode tubes maybe
operated as grounded
1.0 IC
POWER ABSORBED Or OUTPUT TUBE GRID AND BIAS SUPPLY.
-
165
circuit. If a conventional filament transformer
is to be used the cathode tank coil may consist of two parallel heavy conductors (to carry
the high filament current) by-passed at both
the ground end and at the tube socket. The
tuning capacitor is then placed between filament and ground.lt is possible in certain cases
to use two r -f chokes of special design to feed
the filament current to the tubes, with a conventional tank circuit between filament and
ground. Coaxial lines also may be used to
serve both as cathode tank and filament feed
to the tubes for v -h -f and u -h -f work.
-
POWER DELIVERED BV OUTPUT TUBE -
Amplifier
-G
POWER
AMPLIFIER
Showing the relationships between the tube
potentials and currents and the input and
output power of the stage. The approximate
grid impedance also is given.
grid
(cathode
driven) amplifiers by tying the grid and screen
together and operating the tube as a high -u
triode (figure 11). Combined grid and screen
current, however, is a function of tube geometry and may reach destructive values under
conditions of full excitation. Proper division
of excitation between grid and screen should
be as the ratio of the screen -to -grid amplification, which is approximately 5 for tubes
such as the 4 -250A, 4 -400A, etc. The proper
ratio of grid /screen excitation may be achieved by tapping the grid at some point on the
filament choke, as shown. Grid dissipation is
reduced, Lut the overall level of excitation is
increased about 30% over the value required
for simple grounded -grid operation.
The design of such an amplifier stage is
design of a
grounded -grid amplifier stage as far as the
first step is concerned. Then, for the second
step the operating conditions given in figure
10 are applied to the data obtained in the first
step. As an example, take the 304TL stage
previously described. The total power required
of the driver will be (from figure 10) approximately (2700X0.58);1.8) /2 or 141 watts. Of
this 141 watts 27.5 watts (as before) will be
lost as grid dissipation and bias loss and the
balance of 113.5 watts will appear as output.
The total output of the stage will then be approximately 963 watts.
essentially the same as the
4 -200A, 4 -400A,
fFt'
ORIVe
RFC-
RFC
rl
Cathode Tank for
G -G or C -F
Power Amplifier
The cathode tank circuit
for either a grounded-grid
or cathode -follower r -f
power amplifier may be a
conventional tank circuit if the filament transformer for the stage is of the low-capacitance
high- voltage type. Conventional filament transformers, however, will not operate with the
high values of r -f voltage present in such a
FIGURE II
TAPPED FILAMENT CHORE REDUCES EXCESSIVE
GRID DISSIPATION IN G -G CIRCUIT.
RFC--TWOP
CAC.,
TAP
71- e.a
II
WINDINGS OP 191C WIRE, ES TURN!
DIAM. TOTAL LENGTH IS SIR INCHES. GRID
TURNS PROM GROUND CND OP ONE WINDING.
I-
1
vOLTS AT
AMPERE!. (VOLTAGE DROP ACROSS
I.a VOLTS )
RFC Is
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-F
Vacuum
Tube
THE
,'Amplifiers
166
R
8 -6
Class ABI Radio Frequency
Power Amplifiers
»
yll
i
iiiiiis
vis
w
w1i
/j
o
PORTIONOF
EG-tP
Class Aß1 r -f amplifiers operate under
such conditions of bias and excitation that
grid current does not flow over any portion of
the input cycle. This is desirable, since
distortion caused by grid current loading is
absent, and also because the stage is capable
of high power gain. Stage efficiency is about
i 91JJiT1-!ßr
-
MOSTLINRWA,-P
CURVE
.Ì*n
15111111111114
MIN
RADIO
A,
SIGNAL
111
IN
plate current operating angle of
2100 is chosen, as compared to 62% for Class
58% when a
operation.
The level of static (quiescent) plate current
for lowest distortion is quite critical for
Class AB1 tetrode operation. This value is
determined by the tube characteristics, and is
not greatly affected by the circuit parameters
or operating voltages. The maximum d -c plate
potential is therefore limited by the static
dissipation of the tube, since the resting plate
current figure is fixed. The static plate current
of a tetrode tube varies as the 3/2 power of
the screen voltage. For example, raising the
screen voltage from 300 to 500 volts will
double the plate current. The optimum static
plate current for minimum distortion is also
doubled, since the shape of the Eg -Ip curve
does not change.
In actual practice, somewhat lower static
plate current than optimum may be employed
without raising the distortion appreciably,
and values of static plate current of 0.6 to
0.8 of optimum may be safely used, depending
upon the amount of nonlinearity that can be
tolerated.
As with the class B linear stage, the minimum plate voltage swing of the class AB1
amplifier must be kept above the d-c screen
potential to prevent operation in the nonlinear
portion of the characteristic curve. A low value
of screen voltage allows greater r -f plate
voltage swing, resulting in improvement in
plate efficiency of the tube. A balance between plate dissipation, plate efficiency, and
plate voltage swing must be achieved for best
linearity of the amplifier.
B
The 5 -Curve
The perfect linear amplifier delivers a signal
that is a replica of the input signal. Inspection
of the plate characteristic curve of a typical
tube will disclose the tube linearity under
class A operating conditions (figure 12). The
curve is usually of exponential shape, and
the signal distortion is held to a small value
by operating the tube well below its maximum
output, and centering operation over the most
linear portion of the characteristic curve.
Figure 12
CURVE
Eg -Ip
Amplifier
most
operation is confined to
linear portion of characteristic
curve.
The relationship between exciting voltage
in a Class ABI amplifier and the r -f plate
circuit voltage is shown in figure 13. With a
small value of static plate current the lower
portion of the line is curved. Maximum undistorted output is limited by the point on the
line (A) where the instantaneous plate voltage
down to the screen voltage. This "hook" in
the line is caused by current diverted from
the plate to the grid and screen elements of
the tube. The characteristic plot of the usual
linear amplifier takes the shape of an S-curve.
The lower portion of the curve is straightened
out by using the proper value of static plate
current, and the upper portion of the curve is
avoided by limiting minimum plate voltage
swing to a point substantially above the value
of the
screen voltage.
The approximate operacing parameters may be
Linear Amplifier
obtained from the Constant Current curves
(Eg -Ep) or the Eg -Ip curves of the tube in
question. An operating load line is first
approximated. One end of the load line is
determined by the d -c operating voltage of the
tube, and the required static plate current.
As a starting point, let the product of the plate
voltage and current equal the plate dissipation
of the tube. Assuming we have a 4 -400A
tetrode, this end of the load line will fall on
point A (figure 14). Plate power dissipation
is 360 watts (3000V @ 120 ma). The opposite
end of the load line will fall on a point determined by the minimum instantaneous plate voltage, and by the maximum instantaneous plate
current. The minimum plate voltage, for best
linearity should be considerably higher than
Operating Parameters
for the Class ABI
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HANDBOOK
ABI
R.F.
Power Amplifiers
167
therefore the load line cannot cross the Eg =0
line. At the point Ep =600, Eg =0, the maximum
plate current is 580 ma (Point "B ").
Each point the load line crosses a grid voltage axis may be taken as a point for construction of the Eg -Ip curve, just as was done in
figure 22, chapter 6. A constructed curve
shows that the approximate static bias voltage
is -74 volts, which checks closely with point A
of figure 14. In actual practice, the bias voltage is set to hold the actual dissipation slightly
below the maximum figure of the tube.
R F.
E
our
R.F.
E
iN
The single tone power output is:
Figure 13
LINEARITY CURVE OF
TYPICAL TETRODE AMPLIFIER
Emax -Eminx Ipmax. or
4
3000 -600 x .58=348 watts
4
The plate current -angle efficiency factor for
this class of operation is 0.73, and the actual
At point "A" the instantaneous plate
voltage is swinging down to the value
of screen voltage. At point "B" it is
swinging well below the screen and is
approaching the point where saturation,
or plate current limiting takes place.
plate circuit efficiency is:
Np =Emax -Emin x0.73, or
Emax
3000- 600 X 0.73 = 58.4%
3000
The power input to the stage is therefore
the screen voltage. In this case, the screen
voltage is 500, so the minimum plate voltage
excursion should be limited to 600 volts.
Class AB1 operation implies no grid current,
Pox
100 or, 348
Np
58.4
=
595 watts
The plate dissipation is: 595 -348 =247 watts.
.
4k1RtIoN
Pi- network tetrode
TOP VIEW OF A 4 -250A AMPLIFIER
amplifier may be operated Class AB, Class 8, or Class
C by varying potentials
applied to tube. Same general physical and mechanical design applies in each cose.
www.americanradiohistory.com
168
r,
RADIO
THE
-F Vacuum Tube Amplifiers
R
oo.
4 -400A rues
E SCR =
900 VOLTS
r 'S
r I0'
4
t5L-
t
POINT B
w
o
is
SO
00
oC
woo
2000
,000
S
-
ER
.s
.E
.4
m,_
LOAD LINE
.2
POINT
A
o
I
zs
SO
I
VALUE OF
ER MIN OOOV..
11.2 0.3E A
VALUE OF
MAX.
DISSIPATION
(3000 V. X 0.71
A
.300 wArrs)
75
2 nni
hiclre
14
OPERATING PARAMETERS FOR TETRODE LINEAR
AMPLIFIER ARE OBTAINED FROM CONSTANT- CURRENT
CURVES.
It can be seen that the limiting factor for
this class of operation is the static plate dissipation, which is quite a bit higher than the
operating dissipation level. It is possible, at
the expense of a higher level of distortion, to
drop the static plate dissipation and to increase
the screen voltage to obtain greater power output. If the screen voltage is set at 800, and
the bias increased sufficiently to drop the
static plate current to 90 ma, the single tone
d-c plate current may rise to 300 ma, for a
power input of 900 watts. The plate circuit
efficiency is 55.6%, and the power output is
500 watts. Static plate dissipation is 270 watts.
At a screen potential of 500 volts, the maximum screen current is less than 1 ma, and under
certain loading conditions may be negative.
When the screen potential is raised to 800 volts
maximum screen current is 18 ma. The performance of the tube depends upon the voltage
fields set up within the tube by the cathode,
control grid, screen grid, and plate. The quantity
of current flowing in the screen circuit is only
incidental to the fact that the screen is maintained at a positive potential with respect to
the electron stream surrounding it.
The tube will perform as expected so long as
the screen current, in either direction, does
not create undesirable changes in the screen
voltage, or cause excessive screen dissipation.
Good regulation of the screen supply is there-
required. Screen dissipation is highly
responsive to plate loading conditions, and the
plate circuit should always be adjusted so as
to keep the screen current below the maximum
dissipation level as established by the applied
voltage.
G -G Class B Linear
Certain tetrode and pentode
tubes, such as the 6AG7,
Tetrode Amplifier
837, and 803 perform well
as grounded grid class B linear amplifiers. In
this configuration both grids and the suppressor
are grounded, and excitation is applied to the
cathode circuit of the tube. So connected, the
tubes take on characteristics of high -mu triodes.
No bias or screen supplies are required for
this type of operation, and reasonably linear
fore
6AG7
-r
DRIVER
B4-300-700
Figure
15
SIMPLE GROUNDED -GRID
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LINEAR AMPLIFIER
v.
169
HANDBOOK
V2
VI
837
837
500
803
500
803
500
250
250
250
250
VS
V4
V3
837
INPUT
2 War TS
PEAK
.001
2
N
=
.001
5KV
=
Ti
E0
115V
Figure
3
16
-STAGE KILOWATT LINEAR
AMPLIFIER FOR 80 OR 40
METER OPERATION
operation can be had with a very minimum of
circuit components (figure 15). The input impedance of the g -g stage falls between 100
and 250 ohms, eliminating the necessity of
swamping resistors, even though considerable
power is drawn by the cathode circuit of the
g -g stage.
Power gain of a g -g stage varies from approximately 20 when tubes of the 6AG7 type
are used, down to five or six for the 837 and
803 tubes. One or more g -g stages may be
cascaded to provide up to a kilowatt of power,
as illustrated in figure 16.
The input and output circuits of cascaded
g -g stages are in series, and a variation in
load impedance of the output stage reflects
back as a proportional change on the input
circuit. If the first g -g stage is driven by a high
impedance source, such as a tetrode amplifier,
any change in gain will automatically be compensated for. If the gain of V4 -V5 drops, the
input impedance to that stage will rise. This
change will reflect through V2 -V3 so that the
load impedance of VI rises. Since V1 has a
high internal impedance the output voltage
will rise when the load impedance rises. The
increased output voltage will raise the output
voltage of each g -g stage so that the overall
output is nearly up to the initial value before
the drop in gain of V4 -V5.
The tank circuits, therefore, of all g -g stages
must be resonated with low plate voltage and
excitation applied to the tubes. Tuning of one
stage will affect the ocher stages, and the input and coupling of each stage must be adjusted in turn until the proper power limit is
reached.
Operating Dota for
4 -400A Grounded
Grid Linear
Amplifier
ACH
ti
2500
An open frame filament transformer may
be used for TI. Cathode taps are ad-
justed for proper excitation of following
stage.
using a 4 -400A tube for the h.f. region. The
operating characteristics of the amplifier are
summarized in figure 17. It can be noted that
unusually low screen voltage is used on the
tube. The use of lower screen voltage has the
adverse effect of increasing the driving power,
but at the same time the static plate current
of the stage is decreased and linearity is imimproved. For grounded grid operation of the
4 -400A, a screen voltage of 300 volts (filament
to screen) gives a reasonable compromise between these factors.
OPERATING DATA FOR 4- 400A/4 -250A
G -G. LINEAR AB, AMPLIFIER
(SINGLE CONE)
D
-C SCREEN VOLTAGE
D -C
PLATE VOLTAGE
+300
+3000
+3500
60 MA.
STATIC PLATE CURRENT
D -C
+300
-60
GRID BIAS
67
PEAK CATHODE SWING
V.
V.
60 MA.
-59 V.
113 V.
MINIMUM PLATE VOLTAGE
660
MAXIMUM SIGNAL GRID CURRENT
3.6 MA.
10
MAXIMUM SIGNAL SCREEN CURRENT
.1 MA.
20 MA.
MAXIMUM SIGNAL PLATE CURRENT
195 MA.
267 MA.
MAXIMUM SIGNAL PLATE DISSIPATION
235
235
STATIC PLATE DISSIPATION
160 W.
210W.
GRID DRIVING POWER
0.63
3.4
FEEDTHRU POWER
6.55 W.
,s.e
V.
W.
W.
S00 V.
MA.
W.
W.
W.
POWER OUTPUT
(MAXIMUM)
350
W.
700
W.
POWER INPUT
(MAX /MUM)
565
W.
935
W.
Experiments have been
conducted by Collins Radio Co. on a grounded grid
linear amplifier stage
www.americanradiohistory.com
Figure
17
CHAPTER NINE
The Oscilloscope
The cathode -ray oscilloscope (also called
oscillograph) is an instrument which permits
visual examination of various electrical phenomena of interest to the electronic engineer.
Instantaneous changes in voltage, current and
phase are observable if they take place slowly enough for the eye to follow, or if they are
periodic for a long enough time so that the eye
can obtain an impression from the screen of
the cathode -ray tube. In addition, the cathode ray oscilloscope may be used to study any
variable (within the limits of its frequency
response characteristic) which can be converted into electrical potentials. This conversion is made possible by the use of some type
of transducer, such as a vibration pickup unit,
pressure pickup unit, photoelectric cell, microphone, or a variable impedance. The use
of such a transducer makes the oscilloscope
a valuable tool in fields other than electronics.
the recipient of signals from two sources: the
vertical and horizontal amplifiers. The operation of the cathode -ray tube itself has been
covered in Chapter 4; the auxiliary circuits
pertaining to the cathode -ray tube will be
covered here.
The Vertical
The incoming signal which is
to be examined is applied to
the terminals marked Vertical
Input and Ground. The Vertical Input terminal
is connected through capacitor CE (figure 2) so
that the a-c component of the input signal appears across the vertical amplifier gain control potentiometer, R,. Thus the magnitude of
the incoming signal may be controlled to provide the desired deflection on the screen of
Amplifier
To
'ERTICAL
NF T
9 -1
A
Typical Cathode -Ray
Oscilloscope
FLIP.
INTENSITY
MOD.
ATE
ro HORIZONTAL
- -So --
DEFLECTION
EAT.
T,
RE
GND
For the purpose of analysis, the operation
of a simple oscilloscope will be described.
The Du Mont type 274-A unit is a fit instrument for such a description. The block diagram
of the 274 -A is shown in figure 1. The electron beam of the cathode -ray tube can be moved
O
-- -rsa - - - -
/
NoNIZONTAL/ SWEEPS
IUT
/
R-R}DIR7
111111000111111
AM.
vertically or horizontally, or the vertical and
-
DIR
1NFVT
Figure
BLOCK DIAGRAM, TYPE 274 -A
CATHODE -RAY OSCILLOSCOPE
horizontal movements may be combined to produce composite patterns on the tube screen.
As shown in figure 1, the cathode -ray tube is
1
170
www.americanradiohistory.com
SI
STNC.
Time
Base
Generator
171
CI
0.25 LF
INPUT
6AC7
O--{
OUTPUT
VERT. AMP.
Ra
CONTROL
11A
RT.15aK
R30
ee M
Cz
R2
5100
1K
Figure
2
TYPICAL AMPLIFIER SCHEMATIC
Figure
the cathode -ray tube. Also, as shown in figure
1, S, has been incorporated to by -pass the
vertical amplifier and capacitively couple the
input signal directly to the vertical deflection
plate
if
so
desired.
In figure 2, V, is a 6AC7 pentode tube which
is used as the vertical amplifier. As the signal variations appear on the grid of
variations in the plate current of V, will take place.
Thus signal variations will appear in opposite
phase and greatly amplified across the plate
resistor, R,. Capacitor C, has been added across R, in the cathode circuit of V, to flatten
the frequency response of the amplifier at the
high frequencies. This capacitor because of
its low value has very little effect at low input frequencies, but operates more effectively
as the frequency of the signal increases. The
amplified signal delivered by V, is now applied through the second half of switch S, and
capacitor C, to the free vertical deflection
plate of the cathode -ray tube (figure 3).
V
The circuit of the horizontal
amplifier and the circuit of
the vertical amplifier, described in the above paragraph, are similar. A
switch in the input circuit makes provision for
the input from the Horizontal Input terminals
to be capacitively coupled to the grid of the
horizontal amplifier or to the free horizontal
deflection plate thus by- passing the amplifier,
or for the output of the sweep generator to be
capacitively coupled to the amplifier, as shown
in figure 1.
The Horizontal
Amplifier
The Time Bose
Generator
e l e c t r i c al
wave forms by the use of a
cathode -ray tube frequently
3
SCHEMATIC OF CATHODE -RAY TUBE
CIRCUITS
A 5BPIA cathode -ray tube is used in this
instrument. As shown, the necessary potentials for operating this tube are obtained
from a voltage divider mode up of resistors
R21 through R26 inclusive. The intensity of
the beam is adjusted by moving the contact
on R21. This adjusts the potential on the
cathode more or less negative with respect
to the grid which is operated at the full negative voltage -1200 volts. Focusing to the
desired sharpness is accomplished by adjusting the contact on R23 to provide the
correct potential for anode no. 1. Interdependency between the focus and the, intensity controls is inherent in all electrostatically focused cathode-ray tubes. In short,
there is an optimum setting of the focus control for every setting of the intensity control. The second anode of the 5BP IA is operated at ground potential in this instrument.
Also one of each pair of deflection plates is
operated at ground potential.
The cathode is operated at a high negative
potential (approximately 1200 volts) so that
the total overall accelerating voltage of this
tube is regarded os 1200 volts since the second anode is operated at ground potential.
The vertical and horizontal positioning controls which are connected to their respective
deflection plates are capable of supplying
either a positive or negative d -c potential
to the deflection plates. This permits the
spot to be positioned at any desired place
on the entire screen.
Investigation of
requires that some means be readily available
to determine the variation in these wave forms
with respect to time. When such a time base
is required, the patterns presented on the cathode -ray tube screen show the variation in amplitude of the input signal with respect to
time. Such an arrangement is made possible
by the inclusion in the oscilloscope of a Time
Base- Generator. The function of this generator is to move the spot across the screen at a
constant rate from left to right between two
selected points, to return the spot almost instantaneously to its original position, and to
www.americanradiohistory.com
172
The
Oscilloscope
884 tube, the tube
THE
RADIO
will ionize
(or fire) at a
specific plate voltage.
Capacitors C3.-C24 are selectively connected
in parallel with the 884 tube. Resistor
limits the peak current drain of the gas triode.
The plate voltage on this tube is obtained
through resistors R22,
and R11. The voltage
applied to the plate of the 884 tube cannot reach
the power supply voltage because of the charging effect this voltage has upon the capacitor
which is connected across the tube. This capacitor charges until the plate voltage becomes
high enough to ionize the gas in the tube. At
this time, the 884 tube starts to conduct and
the capacitor discharges through the tube until
its voltage falls to the extinction potential of
the tube. When the tube stops conducting, the
capacitor voltage builds up until the tube fires
again. As this action continues, it results in
the sawtooth wave form of figure 4 appearing
at the junction of
and R77.
R
Figure
SAWTOOTH
R
4
WAVE FORM
repeat this procedure at a specified rate. This
action is accomplished by the voltage output
from the time base (sweep) generator. The
rate at which this voltage repeats the cycle
of sweeping the spot across the screen is referred to as the sweep frequency. The sweep
voltage necessary to produce the motion described above must be of a sawtooth waveform, such as that shown in figure 4.
The sweep occurs as the voltage varies
from A to B, and the return trace as the voltage varies from B to C. If A-B is a straight
line, the sweep generated by this voltage will
be linear. It should be realized that the sawtooth sweep signal is only used to plot variations in the vertical axis signal with respect
to time. Specialized studies have made necessary the use of sweep signals of various
shapes which are introduced from an external
source through the Horizontal Input terminals.
R
Synchronization
the sweep generator may be
synchronized from the vertical amplifier or
from an external source. The switch S, shown
in figure 5 is mounted on the front panel to be
easily accessible to the operator.
If no synchronizing voltage is applied, the
discharge tube will begin to conduct when the
plate potential reaches the value of F.t (Firing
Potential). When this breakdown takes place
and the tube begins to conduct, the capacitor
is discharged rapidly through the tube, and the
plate voltage decreases until it reaches the
extinction potential E1. At this point conduction ceases, and the plate potential rises slowly as the capacitor begins to charge through
R7, and R25. The plate potential will again
reach a point of conduction and the circuit
will start a new cycle. The rapidity of the
plate voltage rise is dependent upon the circuit
constants R77 R25, and the capacitor selected,
The sawtooth voltage necessary to obtain the linear time
base is generated by the circuit of figure 5, which operates as follows:
A type 884 gas triode (V3) is used for the
sweep generator tube. This tube contains an
inert gas which ionizes when the voltage between the cathode and the plate reaches a certain value. The ionizing voltage depends upon
the bias voltage of the tube, which is determined by the voltage divider resistors R12 -R17.
With a specific negative bias applied to the
The Sawtooth
Generator
-
C10,0
5
ur.
GÉNECN1T011
CII,O.IUr
C 12, 03
C
T
OUTPUT
ur
1001212!
CN,22012121
SIGNAL TO
DEFLECTION PLATE
Rr
ISO
470
RE
6AC7
E
Provision has been made so
0
nTERr.A
Ree
Su
EAT.
FINE MM
CONTROL
SYNC
R a100A
1200
Figure
SCHEMATIC
OF
TO
St
5
SWEEP
GENERATOR
www.americanradiohistory.com
HANDBOOK
The
Oscilloscope
173
Eb +
EPVSEg
EP+
STATIC CONTROL
f
CHARACTERISTIC
Er
- --
Irl
I
FREE RUN
HING PERIOD
t+
D.C.GRID
BIAS
FIRING POTENTIAL
(D.C. BIAS)
Eex
FIRING
011
POTENTIA
WITH SYNC.
SIGNAL
SYNCHRONIZE
PERIOD
EXTINCTION
POTENTIAL
-EJ
SYNC. SIGNAL
APPLIED TO GRID
Figure 6
ANALYSIS OF SYNCHRONIZATION OF
C10 -C14, as well as the supply voltage
The exact relationship is given by:
Ec
Eb(1_erct
Eb
TIME -BASE
GENERATOR
time the plate potential rises to a sufficient
value, so that the sweep recurs at the same
or an integral sub-multiple of the synchronizing signal rate. This is illustrated in figure 6.
)
Figure - shows the power supply to be made up of two definite sections: a low voltage positive supply
which provides power for operating the amplifiers, the sweep generator, and the positioning
circuits of the cathode -ray tube; and the high
voltage negative supply which provides the
potentials necessary for operating the various
Power Supply
Where E,----=Capacitor voltage at time t
Eb =Supply voltage (B+ supply - cathode
bias)
Er---Firing potential or potential at which
time-base gas triode fires
Ex =Extinction potential or potential at
which time-base gas triode ceases
to conduct
e= Base of natural logarithms
t= Time in seconds
r =Resistance
c =Capacity in
in ohms
(R=T + Rzs)
farads (C10, II,
12,
,,, or
14)
The frequency of oscillation will be approximately:
1
f=rc Et-Ex
Ebl
Under this condition (no synchronizing signal applied) the oscillator is said to be /ree
running.
When a positive synchronizing voltage is
applied to the grid, the firing potential of the
tube is reduced. The tube therefore ionizes at
a lower plate potential than when no grid signal is applied. Thus the applied snychronizing voltage fires the gas -filled triode each
Figure 7
SCHEMATIC OF POWER SUPPLY
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174
The Oscilloscope
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www.americanradiohistory.com
HANDBOOK
TIME
Display
of
Waveforms
175
-+
4 SEC.
IFigure
9
PROJECTION
DRAWING OF A SINEWAVE
APPLIED TO THE VERTICAL AXIS AND A
SAWTOOTH WAVE OF THE SAME FRE-
QUENCY APPLIED SIMULTANEOUSLY
THE HORIZONTAL AXIS
ON
electrodes of the cathode -ray tube, and for
certain positioning controls.
The positive low voltage supply consists
of full -wave rectifier (V,), the output of which
is filtered by a capacitor input filter (20 -20 µfd.
and 8 II). It furnishes approximately 400 volts.
The high voltage power supply employs a half
wave rectifier tube, V,. The output of this rectifier is filtered by a resistance -capacitor filter consisting of 0.5 -0.5 pfd. and .18 M. A
voltage divider network attached from the output of this filter obtains the proper operating
potentials for the various electrodes of the
cathode -ray tube. The complete schematic of
the Du Mont 274 -A Oscilloscope is shown in
figure 8.
9 -2
Display of Waveforms
Together with a working knowledge of the
controls of the oscilloscope, an understanding
of how the patterns are traced on the screen
must be obtained for a thorough knowledge of
oscilloscope operation. With this in mind a
careful analysis of two fundamental waveform
patterns is discussed under the following
headings:
a. Patterns plotted against time (using the
sweep generator for horizontal deflection).
b. Lissajous Figures (using a sine wave for
horizontal deflection).
Patterns Plotted
Against Time
A sine wave is typical of
such a pattern and is con-
venient for this study. This
Figure 10
PROJECTION DRAWING SHOWING THE RESULTANT PATTERN WHEN THE FREQUENCY OF THE SAWTOOTH IS ONE -HALF
OF THAT EMPLOYED IN FIGURE 9
amplified by the vertical amplifier
and impressed on the vertical (Y -axis) deflec-
wave is
tion plates of the cathode -ray tube. Simultaneously the sawtooth wave from the time base
generator is amplified and impressed on the
horizontal (X -axis) deflection plates.
The electron beam moves in accordance
with the resultant of the sine and sawtooth
signals. The effect is shown in figure 9 where
the sine and sawtooth waves are graphically
represented on time and voltage axes. Points
on the two waves that occur simultaneously
are numbered similarly. For example, point 2
on the sine wave and point 2 on the sawtooth
wave occur at the same instant. Therefore the
position of the beam at instant 2 is the resultant of the voltages on the horizontal and vertical deflection plates at instant 2. Referring
to figure 9, by projecting lines from the two
point 2 positions, the position of the electron
beam at instant 2 can be located. If projections were drawn from every other instantaneous position of each wave to intersect on the
circle representing the tube screen, the intersections of similarly timed projections would
trace out a sine wave.
In summation, figure 9 illustrates the principles involved in producing a sine wave trace
on the screen of a cathode -ray tube. Each intersection of similarly timed projections represents the position of the electron beam acting under the influence of the varying voltage
waveforms on each pair of deflection plates.
Figure 10 shows the effect on the pattern of
decreasing the frequency of the sawtooth
www.americanradiohistory.com
176
The
Oscilloscope
THE
RADIO
B
Figure
12
METHOD OF CALCULATING FREQUENCY
RATIO OF LISSAJOUS FIGURES
Figure 11
PROJECTION DRAWING SHOWING THE RE-SULTANT LISSAJOUS PATTERN WHEN A
SINE WAVE APPLIED TO THE HORIZONTAL AXIS IS THREE TIMES THAT APPLIED TO THE VERTICAL AXIS
wave. Any recurrent waveform plotted against
time can be displayed and analyzed by the
same procedure as used in these examples.
The sine wave problem just illustrated is
typical of the method by which any waveform
can be displayed on the screen of the cathode ray tube. Such waveforms as square wave,
sawtooth wave, and many more irregular recurrent waveforms can be observed by the same
method explained in the preceding paragraphs.
9 -3
Obtaining a Lissalous
Pattern on the screen
Oscilloscope Settings
1.
The horizontal am-
plifier should be discon-
nected from the sweep
oscillator. The signal
to be examined should be connected to the
horizontal amplifier of the oscilloscope.
2. An audio oscillator signal should be connected to the vertical amplifier of the oscilloscope.
3. By adjusting the frequency of the audio
oscillator a stationary pattern should be obtained on the screen of the oscilloscope. It is
not necessary to stop the pattern, but merely
to slow it up enough to count the loops at the
side of the pattern.
4. Count the number of loops which intersect
an imaginary vertical line AB and the number
of loops which intersect the imaginary horizontal line BC as in figure 12. The ratio of
the number of loops which intersect AB is to
Lissajous Figures
Another fundamental pattern is the Lissajous
figure, named after the 19th century French
scientist. This type of pattern is of particular
use in determining the frequency ratio between
two sine wave signals. If one of these signals
is known, the other can be easily calculated
from the pattern made by the two signals upon
the screen of the cathode -ray tube. Common
practice is to connect the known signal to the
horizontal channel and the unknown signal to
the vertical channel.
The presentation of Lissajous figures can
be analyzed by the same method as previously
used for sine wave presentation. A simple example is shown in figure 11. The frequency
ratio of the signal on the horizontal axis to the
signal on the vertical axis is 3 to 1. If the
known signal on the horizontal axis is 60 cycles per second, the signal on the vertical
axis is 20 cycles.
O
RATIO
I
O
I
RATIO
O RATIO 5
Figure 13
OTHER LISSAJOUS PATTERNS
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2I
HANDBOOK
HHASE
Lissajous
DIFFERENCE =O
PHASE DIFFERENCE
190
PHASE DIFFERENCE
PHASE
-5
OIFFERENCE'225
Figure
LISSAJOUS PATTERNS OBTAINED
PHASE DIFFERENCE
Figure 13 shows other examples of Lissa jous figures. In each case the frequency ratio
shown is the frequency ratio of the signal on
the horizontal axis to that on the vertical
ve r ti c
axis.
Phase Differonce Patterns
Coming under the heading of
Lissajous figures is the method
used to determine the phase
difference between signals of the same frequency. The patterns i n vol v e d take on the
form of ellipses with different degrees of ec-
centricity.
The following steps should
be taken to obphase -difference pattern:
1. With no signal input to the oscilloscope,
the spot should be centered on the screen
of the tube.
2. Connect one signal to the vertical amplifier of the oscilloscope, and the other
signal to the horizontal amplifier.
3. Connect a common ground between the
two frequencies under investigation and
the oscilloscope.
.90.
270
177
PHASE DIFFERENCE=135
PHASE DIFFERENCE
315
14
FROM THE
the number of loops which intersect BC as the
frequency of the horizontal signal is to the
frequency of the vertical signal.
tain
PHASE DIFFERENCE
Figures
MAJOR PHASE
DIFFERENCE ANGLES
plifier control is adjusted
(3 inches). Reconnect the signal to the vertical amplifier.
The resulting pattern will give an accurate
picture of the exact phase difference between
the two waves. If these two patterns are exactly the same frequency but different in phase
and maintain that difference, the pattern on
the screen will remain stationary. If, 'however,
one of these frequencies is drifting slightly,
the pattern will drift slowly through 360°. The
phase angles of 0 °, 45 °, 90 °, 135 °, 180 °,
225 °, 270 °, 315° are shown in figure 14.
Each of the eight patterns in figure 14 can
be analyzed separately by the previously used
a
Adjust the vertical amplifier gain so as
to give about 3 inches of deflection on a
5 inch tube, and adjust the calibrate d
scale of the oscilloscope so that the vertical axis of the scale coincides precisely with the vertical deflection of the spot.
5. Remove the signal from the vertical amplifier, being careful not to change the
setting of the vertical gain control.
6. Increase the gain of the horizontal amplifier to give a deflection exactly the
same as that to which the vertical am-
TIME
-
4.
Figure 15
PROJECTION DRAWING SHOWING THE RESULTANT PHASE DIFFERENCE PATTERN
OF TWO SINE WAVES 45° OUT OF PHASE
www.americanradiohistory.com
178
Oscilloscope
The
Y
Y
INTERCEPT =O
/
SINE
Y
THE
MAXIMUM =I
Y
MAXIMUM.
SINE=
MAXIMUM
=
I
'I MA IMU
MAXIMUM= I
SINE e=
_S
So
INTERCEPT'.S
SINEe',
s=150-
projecti ',n method. Figure 15 shows two sine
waves which differ in phase being projected
on to the screen of the cathode -ray tube. These
signals represent a phase difference of 45 °.
It is extremely important: (1) that the spot
has been centered on the screen of the cathode ray tube, (2) that both the horizontal and vertical amplifiers have been adjusted to give
exactly the same gain, and (3) that the calibrated scale be originally set to coincide with
the displacement of the signal along the vertical axis. If the amplifiers of the oscilloscope
are not used for conveying the signal to the
deflection plates of the cathode -ray tube, the
coarse frequency switch should be set to horizontal input direct and the vertical input
MODULATED
Y
'I
DIFFERENCE
Figure 17
MODULATION
I
¿;-:.
Figure 16
EXAMPLES SHOWING THE USE OF THE FORMULA
TRAPEZOIDAL
=
INTERCEPT =.5
s' o
YINTERCEil,
s=so
Y
RADIO
FOR DETERMINATION OF
PHASE
switch to direct and the outputs of the two
signals must be adjusted to result in exactly
the same vertical deflection as horizontal deflection. Once this deflection has been set by
either the oscillator output controls or the amplifier gain controls in the oscillograph, it
should not be changed for the duration of the
measurement.
Determination of
the Phase Angle
The relation commonly used
in determining the phase
angle between signals is:
Y intercept
Sine
9
Y maximum
PATTERN
Figure 18
CARRIER WAVE PATTERN
Figure 19
PROJECTION DRAWING SHOWING TRAPEZOIDAL PATTERN
www.americanradiohistory.com
HANDBOOK
Trapezoidal Pattern
MODULATED
CARRIER
R F.
179
POWER AMPLIFIER
1-0
ANTENNA
TIME
EACH 1M,
1
MODULATOR 500.11ÁF
STAGE
-
/SAW TOOTH
10000 V.
TV CAPACITOR
SWEEP
CRO
LC TUNES
TO OPERATING FREQUENCY
Figure 20
PROJECTION DRAWING SHOWING MODOLATED CARRIER WAVE PATTERN
C
e+
intercept
=
=
Y maximum
=
where
Y
9
phase angle between signals
point where ellipse crosses vertical axis measured in tenths of
inches. (Calibrations on the
calibrated screen)
highest vertical point on ellipse
in tenths of inches
Several examples of the use of the formula are
given in figure 16. In each case the Y intercept and Y maximum are indicated together
with the sine of the angle and the angle itself.
For the operator to observe these various patterns with a single signal source such as the
test signal, there are many types of phase
shifters which can be used. Circuits can be
obtained from a number of radio text books.
The procedure is to connect the original signal to the horizontal channel of the oscilloscope and the signal which has passed through
the phase shifter to the vertical channel of
the oscilloscope, and follow the procedure set
forth in this discussion to observe the various
phase shift patterns.
9 -4
Monitoring Transmitter
Performance with the Oscilloscope
The oscilloscope may be used as an aid for
the proper operation of a radiotelephone transmitter, and may be used as an indicator of the
overall performance of the transmitter output
signal, and as a modulation monitor.
There are two types of patterns
that can serve as indicators, the
trapezoidal pattern (figure 17) and the modu-
Waveforms
NOTE'
IF
L
_
PICKUP IS INSUFFICIENT,
A TUNED CIRCUIT MAY BE USED
AT THE OSCILLOSCOPE AS SHOWN.
R F.
Figure 21
MONITORING CIRCUIT FOR TRAPEZOIDAL MODULATION PATTERN
laced wave pattern (figure 18). The trapezoidal
pattern is presented on the screen by impressing a modulated carrier wave signal on the vertical deflection plates and the s i g n a l t h a t
modulates the carrier wave signal (the modulating signal) on the horizontal deflection
plates. The trapezoidal pattern can be analyzed by the method used previously in analyzing waveforms. Figure 19 shows how the signals cause the electron beam to trace out the
pattern.
The modulated wave pattern is accomplished
by presenting a modulated carrier wave on the
vertical deflection plates and by using the
time -base generator for horizontal deflection.
The modulated wave pattern also can be used
for analyzing waveforms. Figure 20 shows how
the two signals cause the electron beam to
trace out the pattern.
oscilloscope connections for obtaining a trapezoidal pattern are shown in
figure 21. A portion of the audio output of the
transmitter modulator is applied to the horizontal input of the oscilloscope. The vertical
amplifier of the oscilloscope is disconnected,
and a small amount of modulated r -f energy is
coupled directly to the vertical d e f l e c t i o n
plates of the oscilloscope. A small pickup
loop, loosely coupled to the final amplifier
tank circuit and connected to the vertical deThe Trapezoidal
Pattern
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The
Oscilloscope
The
180
T
H E
RADIO
i
T
EMIN
E
MAX
1
TRAPEZOIDAL WAVE PATTERN
Figure
22
Figure
(L ESS THAN 100^; MODULATION)
(100
mula:
Emax
Emax
=
- Emin x
t
Emin
Figure 24
MODULATION)
flection plates by a short length of coaxial
line will suffice. The amount of excitation to
the plates of the oscilloscope may be adjusted
to provide a pattern of convenient size. Upon
modulation of the transmitter, the trapezoidal
pattern will appear. By changing the degree of
modulation of the carrier wave the shape of
the pattern will change. Figures 22 and 23
show the trapezoidal pattern for various degrees of modulation. The percentage of modulation may be determined by the following forModulation percentage
23
(OVER MODULATION)
figure 25. The internal sweep circuit of the
oscilloscope is applied to the horizontal
plates, and the modulated r -f signal is applied
to the vertical plates, as described before. If
desired, the internal sweep circuit may be snychronized with the modulating signal of the
transmitter by applying a small portion of the
modulator output signal to the external sync
post of the oscilloscope. The percentage of
modulation may be determined in the same
fashion as with a trapezoidal pattern. Figures
26, 27 and 28 show the modulated wave pattern for various degrees of modulation.
100
where Emax and Emin are defined as in
figure 22.
An overmodulated signal is shown in figure
9 -5
Receiver -F Alignment
with an Oscilloscope
I
24.
The Modulated
Wove Pattern
R
F.
The oscilloscope connections
for obtaining a modulated
wave pattern are shown in
POWER AMPLIFIER
TO
CRO
ANTENNA
USE INTERNAL
SWEE
a
FROM
MODUL ATOR
LC TUNES TO OP-
ERATING FREQUENCY
Figure 25
MONITORING CIRCUIT FOR
MODULATED WAVE PATTERN
The alignment of the i -f amplifiers of a receiver consists of adjusting all the tuned circuits to resonance at the intermediate frequency and at the same time to permit passage of
a predetermined number of side bands. The
best indication of this adjustment is a resonance curve representing the response of the
i -f circuit to its particular range of frequencies.
As a rule medium and low- priced receivers
use i -f transformers whose bandwidth is about
5 kc. on each side of the fundamental frequency. The response curve of these i -f transformers is shown in figure 29. High fidelity receivers usually contain i -f transformers which
have a broader bandwidth which is usually 10
kc. on each side of the fundamental. The response curve for this type transformer is shown
in figure 30.
Resonance curves such as these can be displayed on the screen of an oscilloscope. For
a complete understanding of the procedure it
is important to know how the resonance curve
is traced.
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HANDBOOK
Receiver Alignment
VV
EMIN
E
181
V\I"
MLY.
1\1\
CARRIER WAVE PATTERN
Figure 27
Figure 26
(100% MODULATION)
(LESS THAN 100% MODULATION)
The Resonance
Curve on the
Screen
To present a resonance curve
on the screen, a frequency modulated signal source must
be
avail a b l
e.
Figure 28
This signal
source is a signal generator whose output is
the fundamental i -f frequency which is frequency- modulated 5 to 10 kc. each side of the
fundamental frequency. A signal generator of
this type generally takes the form of an ordinary signal generator with a rotating motor
driven tuned circuit capacitor, called a uwob-
(OVER MODULATION)
bulator, or its electronic equivalent,
a
react-
ance tube.
The method of presenting a resonance curve
on the screen is to connect the vertical channel of the oscilloscope across the detector
load of the receiver as shown in the detectors
of figure 31 (between point A and ground) and
the time -base generator output to the horizontal channel. In this way the d -c voltage across
the detector load varies with the frequencies
which are passed by the i -f system. Thus, if
the time -base generator is set at the frequency
of rotation of the motor driven capacitor, or
the reactance tube, a pattern resembling figure 32, a double resonance curve, appears on
the screen.
Figure 32 is explained by considering fighalf a rotation of the motor driven
capacitor the frequency increases from 445
kc. to 465 kc., more than covering the range
of frequencies passed by the i -f system.
Therefore, a full resonance curve is presented
on the screen during this half cycle of rotation since only half a cycle of the voltage producing horizontal deflection has transpired.
In the second half of the rotation the motor
ure 33. In
eKC
K
4
KC ecc
Figure 29
FREQUENCY RESPONSE CURVE OF THE
I -F OF A LOW PRICED RECEIVER
ecc
TRIODE DETECTOR
eKc
DIODE DETECTOR
Figure
Figure 30
FREQUENCY RESPONSE OF
HIGH- FIDELITY I -F SYSTEM
31
CONNECTION OF THE OSCILLOSCOPE
ACROSS THE DETECTOR LOAD
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The
182
Figure 32
DOUBLE RESONANCE CURVE
445
KC
455 KC
THE
Oscilloscope
46
455 KC
KC
445 KC
Figure 33
DOUBLE
RESONANCE ACHIEVED BY
COMPLETE ROTATION OF THE MOTOR
DRIVEN CAPACITOR
curve is observed as it sweeps the spot across
the screen from left to right; and it is observed
again as the sine wave sweeps the spot back
again from right to left. Under these conditions the two response curves are superimposed on each other and the high frequency
responses of both curves are at one end and
the low frequency response of both curves is
at the other end. The i -f trimmer capacitors
are adjusted to produce a response curve
which is symmetrical on each side of the fundamental frequency.
When using sawtooth sweep, the two response curves can also be superimposed. If
the sawtooth signal is generated at exactly
twice the frequency of rotation of the motor
driven capacitor, the two resonance curves
will be superimposed (figure 34) if the i -f
transformers are properly tuned. If the two
curves do not coincide the i -f trimmer capacitors should be adjusted. At the point of coincidence the tuning is correct. It should be
pointed out that rarely do the two curves agree
perfectly. As a result, optimum adjustment is
made by making the peaks coincide. This latter procedure is the one generally used in i -f
adjustment. When the two curves coincide, it
is evident that the i -f system responds equally to signals higher and lower than the fundamental i -f frequency.
9 -6
Figure
34
SUPER -POSITION OF RESONANCE CURVES
driven capacitor takes the frequency of the
signal in the reverse order through the range
of frequencies passed by the i -f system. In
this interval the time -base generator sawtooth
waveform completes its cycle, drawing the
electron beam further across the screen and
then returning it to the starting point. Subsequent cycles of the motor driven capacitor and
the sawtooth voltage merely retrace the same
pattern. Since the signal being viewed is applied through the vertical amplifier, the sweep
can be synchronized internally.
Some signal generators, particularly those
employing a reactance tube, provide a sweep
output in the form of a sine wave which is
synchronized to the frequency with which the
reactance tube is swinging the fundamental
frequency through its limits, usually 60 cycles
per second. If such a signal is used for horizontal deflection, it is already synchronized.
Since this signal is a sine wave, the response
RADIO
Single Sideband Applications
Measurement of power output and distortion
are of particular importance in SSB transmitter
adjustment. These measurements are related to
the extent that distortion rises rapidly when
the power amplifier is overloaded. The useable
power output of a SSB transmitter is often defined as the maximum peak envelope power
II11IIIIIIU114ulul
m
IIIIIIIIIIIIIIIIIIII
Figure
SINGLE
35
TONE PRESENTATION
Oscilloscope trace of SSB signal
modulated by single tone (A).
Incomplete carrier supression or
spurious products will show
modulated envelope of (B). The
ratio of supression is:
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S
-
20 log
A
+B
A -B
S.S.B.
HANDBOOK
R
POWER
VER A
-F INPUT
T
R -F
LIFIER
SSB INPUT
VOLTAGE
FOM
TEST
GERMANIUM
DIODE
tpplications
2.5
183
MM
RFC
AUDIO OUTPUT
70 OSCILLOSCOPE
DIVIDER OR
PICMUP COIL
INPUT
ENVELOPE
DETECTOR
Figure 37
SCHEMATIC OF
ENVELOPE DETECTOR
OSCILLOSCOPE
Figure 36
BLOCK DIAGRAM OF
LINEARITY TRACER
obtainable with a specified signal-to- distortion
ratio. The oscilloscope is a useful instrument
for measuring and studying distortion of all
types that may be generated in single sideband
equipment.
Single Tone
When aSSB
transmitter is modu-
laced with a single audio tone,
the r -f output should be a single
radio frequency. If the vertical plates of the
oscilloscope are coupled to the output of the
transmitter, and the horizontal amplifier sweep
is set to a slow rate, the scope presentation
will be as shown in figure 35. If unwanted distortion products or carrier are present, the top
and bottom of the pattern will develop a "ripple" proportional to the degree of spurious
Observations
products.
The linearity tracer is an auxiliary detector to be used with
an oscilloscope for quick observation of amplifier adjustments and parameter variations. This instrument consists of
two SSB envelope detectors the outputs of
which connect to the horizontal and vertical
inputs of an oscilloscope. Figure 36 shows a
block diagram of atypical linearity test set -up.
A two -tone test signal is normally employed
to supply a SSB modulation envelope, but any
modulating signal that provides an envelope
that varies from zero to full amplitude may be
The Linearity
Tracer
used. Speech modulation gives a satisfactory
trace, so that this instrument may be used as
a visual monitor of transmitter linearity. It is
particularly useful for monitoring the signal
level and clearly shows when the amplifier
under observation is overloaded. The linearity
trace will be a straight line regardless of the
envelope shape if the amplifier has no distortion. Overloading causes a sharp break in
the linearity curve. Distortion due to too much
bias is also easily observed and the adjustment
for low distortion can easily be made.
Another feature of the linearity detector is
that the distortion of each individual stage
can be observed. This is helpful in troubleshooting. By connecting the input envelope
detector to the output of the SSB generator,
the overall distortion of the entire r -f circuit
beyond this point is observed. The unit can
also serve as a voltage indicator which is
useful in making tuning adjustments.
The circuit of a typical envelope detector
is shown in figure 37. Two matched germainum
diodes are used as detectors. The detectors
are not linear at low signal levels, but if the
nonlinearity of the two detectors is matched,
the effect of their nonlinearity on the oscilloscope trace is cancelled. The effect of diode
differences is minimized by using a diode load
of 5,000 to 10,000 ohms, as shown. It is important that both detectors operate at approximately the same signal level so that their
differences will cancel more exactly. The
operating level should be 1 -volt or higher.
It is convenient to build the detector in a
small shielded enclosure such as an i -f transformer can fitted with coaxial input and output
connectors. Voltage dividers can be similarly
constructed so that it is easy to insert the desired amount of voltage attenuation from the
various sources. In some cases it is convenient
to use a pickup loop on the end of a short
length of coaxial cable.
The phase shift of the amplifiers in the oscilloscope should be the same and their frequency response should be flat out to at least
twenty times the frequency difference of the
two test tones. Excellent high frequency characteristics are necessary because the rectified
SSB envelope contains harmonics extending
to the limit of the envelope detector's response.
Inadequate frequency response of the vertical
amplifier may cause a little "foot" to appear
on the lower end of the trace, as shown in
figure 38. If it is small, it may be safely neg-
lected.
Another spurious effect often encountered
is a double trace, as shown in figure 39. This
can usually be corrected with an R -C network
placed between one detector and the oscilloscope. The best method of testing the detectors
and the amplifiers is to connect the input of
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184
The
Oscilloscope
OUTPUT
SIGNAL
LEVEL
Figure 38
EFFECT OF INADEQUATE
RESPONSE
OF
VERTICAL
AMPLIFIER
INPUT SIGNAL LEVEL
Figure 41
ORDINATES ON LINEARITY
CURVE
FOR
3RD
ORDER
DISTORTION EQUATION
Figure 39
DOUBLE TRACE
CAUSED BY PHASE
SHIFT
the envelope detectors in parallel. A perfectly
straight line trace will result when everything
is working properly. One detector is then connected to the other r -f source through a voltage
divider adjusted so that no appreciable change
in the setting of the oscilloscope amplifier
controls is required. Figure 40 illustrates some
typical linearity traces. Trace A is caused by
inadequate static plate current in class A or
class B amplifiers or a mixer stage. To regain
linearity, the grid bias of the stage should be
reduced, the screen voltage should be raised,
or the signal level should be decreased. Trace
B is a result of poor grid circuit regulation
when grid current is drawn, or a result of non-
linear plate characteristics of the amplifier
tube at large plate swings. More grid swamping
should be used, or the exciting signal should
be reduced. A combination of the effects of A
and B are shown in Trace C. Trace D illustrates
amplifier overloading. The exciting signal
should be reduced.
A means of estimating the distortion level
observed is quite useful. The first and third
order distortion components may be derived by
an equation that will give the approximate
signal -to- distortion level ratio of a two tone
test signal, operating on a given linearity curve.
Figure 41 shows a linearity curve with two
ordinates erected at half and full peak input
signal level. The length of the ordinates et
and e2 may be scaled and used in the following
equation:
Signal -to- distortion ratio in db =20 log 8 e t -e2
TYPICAL LINEARITY TRACES
Figure 40
TYPICAL LINEARITY
TRACES
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2
el -e2
CHAPTER TEN
Special Vacuum Tube Circuits
A whole new concept of vacuum tube applications has been developed in recent years.
No longer are vacuum tubes chained to the
field of communication. This chapter is devoted to some of the more common circuits encountered in industrial and military applications of the vacuum tube.
10 -1
The characteristics of a
diode tube are such that the
tube conducts only when the plate is at a positive potential with respect to the cathode. A
positive potential may be placed on the cathode, but the tube will not conduct until the
voltage on the plate rises above an equally
positive value. As the plate becomes more
positive with respect to the cathode, the diode
conducts and passes that portion of the wave
that is more positive than the cathode voltage.
Diodes may be used as either series or parallel limiters, as shown in figure 1. A diode may
be so biased that only a certain portion of the
positive or negative cycle is removed.
Diode Limiters
Limiting Circuits
The term limiting refers to the removal or
suppression by electronic means of the extremities of an electronic signal. Circuits
which perform this function are referred to as
limiters or clippers. Limiters are useful in
wave-shaping circuits where it is desirable to
square off the extremities of the applied signal. A sine wave may be applied to a limiter
circuit to produce a rectangular wave. A
peaked wave may be applied to a limiter circuit to eliminate either the positive or negative peaks from the output. Limiter circuits
are employed in FM receivers where it is necessary to limit the amplitude of the signal applied to the detector. Limiters may be used to
reduce automobile ignition noise in short -wave
receivers, or to maintain a high average level
of modulation in a transmitter. They may also
be used as protective devices to limit input
signals to special circùits.
An audio peak clipper consisting
of two diode limiters may be used
to limit the amplitude of an audio signal to a predetermined value to provide
a high average level of modulation without
danger of overmodulation. An effective limiter
for this service is the series -diode gate clipper. A circuit of this clipper is shown in figure 2. The audio signal to be clipped is coupled to the clipper through C,. R, and R2 are
the clipper input and output load resistors.
The clipper plates are tied together and are
connected to the clipping level control, R.,
through the series resistor, R3. R. acts as a
voltage divider between the high voltage supply and ground. The exact point at which clipAudio Peak
Limiting
185
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Special Vacuum Tube
186
e iN
Circuits
THE
RADIO
e OUT
E
IAA'
E
e
IN
=
VOLTAGE DROP
ACROSS DIODE
E= VOLTAGE DROP
ACROSS DIODE
e OUT
PIN
e OVT
E
E
lTV i
VT
-A-21
ear
Figure
e OUT
1
VARIOUS DIODE LIMITING CIRCUITS
Series diodes limiting positive and negative peaks are shown in A and
ing positive and negative peaks are shown in C and D. Parallel diodes 8. Parallel diodes limitlimiting above and below
ground are shown in E and F. Parallel diode limiters which pass
negative and positive peaks
are shown in G and H.
ping will occur is set by R,, which controls the
positive potential applied to the diode plates.
Under static conditions, a d -c voltage is obtained from R4 and applied through R, to both
plates of the 6AL5 tube. Current flows through
R,,
and divides through the two diode
sections of the 6AL5 and the two load resistors, R, and Rr. All parts of the clipper circuit
are maintained at a positive potential above
ground. The voltage drop between the plate
and cathode of each diode is very small compared to the drop across the 300,000 -ohm resistor (R,) in series with the diode plates.
The plate and cathode of each diode are therefore maintained at approximately equal potentials as long as there is plate current flow.
Clipping does not occur until the peak audio
input voltage reaches a value greater than the
static voltages at the plates of the diode.
R
Assume that R4 has been set to a point that
will give 4 volts at the plates of the 6AL5.
When the peak audio input voltage is less than
4 volts, both halves of the tube conduct at all
times. As long as the tube conducts, its resistance is very low compared with the plate
resistor R,. Whenever a voltage change occurs
across input resistor
the voltage at all of
the tube elements increases or decreases by
the same amount as the input voltage change,
and the voltage drop across R, changes by an
equal amount. As long as the peak input voltage is less than 4 volts, the 6AL5 acts merely
as a conductor, and the output cathode is permitted to follow all voltage changes at the input cathode.
If, under static conditions, 4 volts appear at
the diode plates, then twice this voltage (8
volts) will appear if one of the diode circuits
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R
Clamping Circuits
HANDBOOK
Ra
6AL5
CLIPPING
LEVEL
300K
CONTROL
C2
Ci
0.1
0.1
e
e IN
OUT
R
00
m
R2
R
zoom
eIN
E
Bt
E
200K
Figure
2
THE SERIES -DIODE GATE CLIPPER FOR
AUDIO PEAK LIMITING
is opened, removing its d -c load from the circuit. As long as only one of the diodes continues to conduct, the voltage at the diode
plates cannot rise above twice the voltage selected by R. In this example, the voltage cannot rise above 8 volts. Now, if the input audio
voltage applied through C, is increased to any
peak value between zero and plus 4 volts, the
first cathode of the 6AL5 will increase in voltage by the same amount to the proper value between 4 and 8 volts. The other tube elements
will assume the same potential as the first
cathode. However, the 6AL5 plates cannot increase more than 4 volts above their original
4 -volt static level. When the input voltage to
the first cathode of the 6AL5 increases to
more than plus 4 volts, the cathode potential
increases to more than 8 volts. Since the plate
circuit potential remains at 8 volts, the first
diode section ceases to conduct until the input voltage across R, drops below 4 volts.
When the input voltage swings in a negative
direction, it will subtract from the 4 -volt drop
across R, and decrease the voltage on the input cathode by an amount equal to the input
voltage. The plates and the output cathode will
follow the voltage level at the input cathode
as long as the input voltage does not swing
below minus 4 volts. If the input voltage does
not change more than 4 volts in a negative
direction, the plates of the 6AL5 will also become negative. The potential at the output
cathode will follow the input cathode voltage
and decrease from its normal value of 4 volts
until it reaches zero potential. As the input
cathode voltage decreases to less than zero,
e
IVENPOSITIVE
A
WNEDNGRIDOSEDR
Figure 3
LIMITING CIRCUIT
GRID
the plates will follow. however, the output
will stop at zero
cathode, grounded through
potential as the plate becomes negative. Conduction through the second diode is impossible
under these conditions. The output cathode
remains at zero potential until the voltage at
the input cathode swings back to zero.
The voltage developed across output resistor R2 follows the input voltage variations as
long as the input voltage does not swing to a
peak value greater than the static voltage at
the diode plates, determined by R. Effective
clipping may thus be obtained at any desired
R
level.
The square-topped audio waves generated
this clipper are high in harmonic content,
but these higher order harmonics may be greatby
ly reduced by a low -level speech filter.
Grid Limiters
A
triode grid limiter is shown
in figure 3. On
positive peaks
of the input signal, the triode grid attempts to
swing positive, and the grid- cathode resistance drops to a value on the order of 1000
ohms or so. The voltage drop across R (usually of the order of I megohm) is large compared to the grid-cathode drop, and the resulting limiting action removes the top part of the
positive input wave.
Clamping Circuits
10 -2
A circuit which holds either amplitude extreme of a waveform to a given reference level
e OUT
IN
187
eiN
eouT
DIODE CONDUCTS
OA POSITIVE CLAMPING CIRCUIT
SIMPLE POSITIVE
©
NEGATIVE CLAMPING CIRCUIT
Figure 4
AND NEGATIVE CLAMPING CIRCUITS
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188
e
Special Vacuum Tube Circuits
r
-,
L-
J
RADIO
THE
¡DEFLECTION
COIL
IN
-100Y
CI CHARGE PATH
NEGATIVE
PLOYED IN
Figure 5
CLAMPING CIRCUIT EMELECTROMAGNETIC SWEEP
C2 DISCHARGE PATH
Figure 7
THE CHARGE AND DISCHARGE PATHS
IN FREE -RUNNING MULTIVIBRATOR OF
FIGURE 6
SYSTEM
is repeated and therefore is "jittery." If a
clamping circuit is placed between the sweep
amplifier and the deflection element, the start
of the sweep can be regulated by adjusting the
d -c voltage applied to the clamping tube (fig-
B+
ure 5).
Multivibrators
10-3
Figure 6
BASIC MULTIVIBRATOR CIRCUIT
The multivibrator, or relaxation oscillator,
is used for the generation of nonsinusoidal
waveforms. The output is rich in harmonics,
but the inherent frequency stability is poor.
The multivibrator may be stabilized by the
introduction of synchronizing voltages of harmonic or subharmonic frequency.
In its simplest form, the multivibrator is a
simple two -stage resistance -capacitance coupled amplifier with the output of the second
stage coupled through a capacitor to the grid
of the first tube, as shown in figure 6. Since
the output of the second stage is of the proper
polarity to reinforce the input signal applied
to the first tube, oscillations can readily take
place, started by thermal agitation noise and
of potential is called a clamping circuit or a
d-c restorer. Clamping circuits are used after
RC cpupling circuits where the wave f o r m
swing is required to be either above or below
the reference voltage, instead of alternating
on both sides of it (figure 4). Clamping circuits are usually encountered in oscilloscope
sweep circuits. If the sweep voltage does not
always start from the same reference point,
the trace on the screen does not begin at the
same point on the screen each time the sweep
B.
B.
NIP
///
SYNCHNONIZING
SIGNAL
B
DIRECT- COUPLED CATHODE
MULTI VIBRATOR
ELECTRON-COUPLED
MULTIVIBRATOR
Figure
©
MULTI VIBRATOR WITH SINE -WAVE
SYNCHRONIZING SIGNAL APPLIED
TO ONE TUBE
8
VARIOUS FORMS OF MULTIVIBRATOR CIRCUITS
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-
Multivibrators
189
PULSE
OUTPUT
ONE -SHOT MULTIVIBRATOR
BASIC ECCLES-JORDAN TRIGGER
CIRCUIT
Figure
9
ECCLES -JORDAN MULTI VIBRATOR CIRCUITS
)
miscellaneous tube noise. Oscillation is maintained by the process of building up and discharging the store of energy in the grid coupling capacitors of the two tubes. The charging and discharging paths are shown in figure
7. Various forms of multivibrators are shown
in figure 8.
The output of a multivibrator may be used
as a source of square waves, as an electronic
switch, or as a means of obtaining frequency
division. Submultiple frequencies as low as
one -tenth of the injected synchronizing frequency may easily be obtained.
The Eccles- Jordan
Circuit
The Eccles -Jordan trigger
circuit is shown in figure
9A. This is not a true mul-
tivibrator, but rather a circuit that possesses
two conditions of stable equilibrium. One condition is when V, is conducting and V2 is cutoff; the other when V2 is conducting and V, is
cutoff. The circuit remains in one or the other
of these two stable conditions with no change
in operating potentials until some external
action occurs which causes the nonconducting
tube to conduct. The tubes then reverse their
functions and remain in the new condition as
long as no plate current flows in the cutoff
tube. This type of circuit is known as a flip -
flop circuit.
Figure 9B illustrates a modified Eccles Jordan circuit which accomplishes a complete
cycle when triggered with a positive pulse.
Such a circuit is called a one -shot multivibrator. For initial action, V, is cutoff and V2 is
conducting. A large positive pulse applied to
the grid of V, causes this tube to conduct, and
the voltage at its plate decreases by virtue of
the IR drop through R3. Capacitor C2 is charged
rapidly by this abrupt change in V, plate voltage, and V, becomes cutoff while V, conducts.
This condition exists until C2 discharges, allowing V2 to conduct, raising the cathode bias
of V, until it is once again cutoff.
A direct, cathode -coupled multivibrator is
shown in figure 8A. RK is a common cathode
resistor for the two tubes, and coupling takes
place across this resistor. It is impossible for
a tube in this circuit to completely cutoff the
other tube, and a circuit of this type is called
a free- running multivibrator in which the condition of one tube temporarily cuts off the
other.
RF
PULSE
RF
RF
PULSE
PULSE
nns
nnl
'1
eoUT
nnl
CUTOFF
TIME
C ouT
I_
Figure 10
BLOCKING
TIME
TIMG
MNI3CIC04E
CUTOFF
TIME
SINGLE -SWING
CUTOFF
OSCILLATOR
Figure
11
HARTLEY OSCILLATOR USED AS BLOCKING
OSCILLATOR BY PROPER CHOICE OF R, -C,
190
Special Vacuum
eIN
Tube
Circuits
e
Pour
POSITIVE COUNTING CIRCUIT
THE
eouT
NEGATIVE COUNTING CIRC,
POSITIVE
'.ETEA
Figure
POSITIVE AND NEGATIVE
10 -4
The Blocking Oscillator
A blocking oscillator is any oscillator which
cuts itself off after one or more cycles caused
by the accumulation of a negative charge on
the grid capacitor. This negative charge may
gradually be drained off through the grid resistor of the tube, allowing the circuit to oscillate once again. The process is repeated
and the tube becomes an intermittent oscillator. The rate of such an occurance is determined by the R-C time constant of the grid circuit. A single -swing blocking oscillator is
shown in figure 10, wherein the tube is cutoff
before the completion of one cycle. The tube
produces single pulses of energy, the time
between the pulses being regulated by the
discharge time of the grid R -C network. The
self-pulsing blocking oscillator is shown in
figure 11, and is used to produce pulses
of r-f energy, the number of pulses being de-
termined by the timing network in the grid circuit of the oscillator. The rate at which these
pulses occur is known as the pulse -repetition
frequency, or p.r. /.
10 -5
ADIO
IN
..
NG
)N
CIRCUIT WITH
12
COUNTING CIRCUITS
ing units to be counted, and produces a- voltage that is proportional to the frequency of
the pulses. A counting circuit may be used in
conjunction with a blocking oscillator to produce a trigger pulse which is a submultiple of
of the frequency of the applied pulse. Either
positive or negative pulses may be counted.
A positive counting circuit is shown in figure
12A, and a negative counting circuit is shown
in figure 12B. The positive counter allows a
certain amount of current to flow through R,
each time a pulse is applied to C,.
The positive pulse charges
and makes
the plate of V, positive with respect to its
cathode. V, conducts until the exciting pulse
passes. C, is then discharged by
and the
circuit is ready to accept another pulse. The
average current flowing through R, increases
as the pulse- repetition frequency increases,
and decreases as the p.r.f. decreases.
By reversing the diode connection s, as
shown in figure 12B, the circuit is made to
C
V
respond to negative pulses. In this circuit, an
increase in the p.r.f. causes a decrease in the
average current flowing through
which is
opposite to the effect in the positive counter.
R
Counting Circuits
A counting circuit, or frequency divider is
one which receives uniform pulses, represent-
e
Vs
feIN
3
e OUTy
Figure
Figure 13
STEP -BY -STEP COUNTING CIRCUIT
14
The step -by -step counter used to trigger a
blocking oscillator. The blocking oscillator
serves as a frequency divider.
www.americanradiohistory.com
HANDBOOK
R
-C
Oscillators
191
4
C2
CI
!00'1.
C
VI
R=LP
RB
LP
R4. 4
a1
xCI =R2
THE
WATT, 110
X
V
LAMP BULB
Ca
Figure 15
WIEN- BRIDGE AUDIO OSCILLATOR
A step- counter is similar to the circuits
discussed, except that a capacitor which is
large compared to C, replaces the diode load
resistor. The charge of this condenser is increased during the time of each pulse, producing a step voltage across the output (figure
13). A blocking oscillator may be connected
to a step- counter, as shown in figure 14. The
oscillator is triggered into operation when the
voltage across C, reaches a point sufficiently
positive to raise the grid of V, above cutoff.
Circuit parameters may be chosen so that a
count division up to 1/20 may be obtained
with reliability.
10 -6
Resistance -Capacity
Oscillators
In an R -C oscillator, the frequency is determined by a resistance capacity network that
provides regenerative coupling between the
output and input of a feedback amplifier. No
use is made of a tank circuit consisting of inductance and capacitance to control the frequency of oscillation.
The Wien - Bridge oscillator employs a Wien
network in the R -C feedback circuit and is
shown in figure 15. Tube V, is the oscillator
tube, and tube V, is an amplifier and phase inverter tube. Since the feedback voltage
through C4 produced by V, is in phase with the
input circuit of V, at all frequencies, oscillation is maintained by voltages of any frequency that exist in the circuit. The bridge circuit
is used, then, to eliminate feedback voltages
of all frequencies except the single frequency
desired at the output of the oscillator. The
bridge allows a voltage of only one frequency
to be effective in the circuit because of the
degeneration and phase shift provided by this
Figure 16
PHASE -SHIFT OSCILLATOR
THE
circuit. The frequency at which oscillation
occurs is:
f-
1
,
2n R, C,
when
R,xC,=R,xC,
Lp is used as the cathode resistor
thermal stabilizer of the oscillator
amplitude. The variation of the resistance
with respect to current of the lamp bulb holds
the oscillator output voltage at a nearly constant amplitude.
The phase -shi /t oscillator shown in figure
16 is a single tube oscillator using a three
section phase shift network. Each section of
the network produces a phase shift in proportion to the frequency of the signal that passes
through it. For oscillations to be produced,
the signal from the plate of the tube must be
shifted 180 °. Three successive phase shifts
of 60° accomplish this, and the frequency of
oscillation is determined by this phase shift A high -mu triode or a pentode must be used in
this circuit. In order to increase the frequency
of oscillation, either the resistance or the
capacitance must be decreased.
A lamp
of V, as
THE
a
Figure 17
BRIDGE -TYPE PHASE -SHIFT
OSCILLATOR
www.americanradiohistory.com
192
Special
Vacuum
Tube
Circuits
THE
RADIO
0-FREQ.
OF
OSCILLATION
NEC F/B =POS F/B
-NOTCHFREQUENCY
F-
NEGATIVE
I
2?RC
FEEDBACK
WHERE
(LOOP
C=1/Ct C2
2)
POSITIVE
FEEDBACK
(LOOP r)
f
rFREQ
OF
OSCILLATION
PHASE
SHIFT'0
"NOTCH NETWORK
Figure 19
BRIDGE -T FEEDBACK
LOOP CIRCUITS
Figure 18
THE NBS BRIDGE -T
OSCILLATOR CIRCUIT AS USED
IN THE HEATH AG -9 AUDIO
GENERATOR
A bridge -type phase shill oscillator is
shown in figure 17. The bridge is so proportioned that at only one frequency is the phase
shift through the bridge 180 °. Voltages of other
frequencies are fed back to the grid of the tube
out of phase with the existing grid signal, and
are cancelled by being amplified out of phase.
The NBS Bridge -T oscillator developed by
the National Bureau of Standards consists of
a two stage amplifier having two feedback
loops, as shown in figure 18. Loop 1 consists
of a regenerative cathode-to- cathode loop, consisting of Lp, and C3, The bulb regulates the
positive feedback, and tends to stabilize the
output of the oscillator, much as in the manner of the Wien circuit. Loop 2 consists of a
grid -cathode degenerative circuit, containing
the bridge -T. Oscillation will occur at the
null frequency of the bridge, at which frequency the bridge allows minimum degeneration
in loop 2 (figure 19).
Oscillation will occur at the null
frequency of the bridge, at which
frequency
the
bridge allows
minimum degeneration in loop 2.
effect system. The furnace (F) raises the
room temperature (T) to a predetermined value
at which point the sensing thermostat (TAI)
reduces the fuel flow to the furnace. When the
room temperature drops below the predetermined value the fuel flow is increased by the
and
thermostat control. An interdependent control
system is created by this arrangement: the
room temperature depends upon the thermostat
action, and the thermostat action depends upon
the room temperature. This sequence of events
may be termed a closed loop feedback system.
ROOM
FURNACE
TEMPERATURE
(F)
(T)
FEEDBACK
(ERROR SIGNAL)
10 -7
Feedback
discus
Feedback amplifiers have been
sed
in Chapter 6, section 15 of this Handbook. A
more general use of feedback is in automatic
control and regulating systems. Mechanical
feedback has been used for many years in such
forms as engine speed governors and steering
servo engines on ships.
A simple feedback system for temperature
control is shown in figure 20. This is a cause
Figure 20
SIMPLE CLOSED LOOP
FEEDBACK SYSTEM
Room temperature (T) controls
fuel supply to furnace (F) by feedloop through Thermostat
back
(TH) control.
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Feedback
HANDBOOK
INPUT SIGNAL
¡OUTPUT SIGNAL
PHASE SHIFT
OF SYSTEM
I!
TIME
OUTPUT SIGNAL
FEEDBACK SIGNAL
NO PHASE SHIFT
.. .
I
A
FEEDBACK
SIGNAL
WITH 180
PHASE
SHIFT
TIME
Figure
21
PHASE SHIFT OF ERROR
SIGNAL MAY CAUSE OSCILLATION INCLOSED LOOP SYSTEM
To prevent oscillation, the gain
of the feedback loop must be
less than unity when the phase
shift of the system reaches 180
degrees.
Error Cancellation
A feedback
control system
is dependent upon a degree
of error in the output signal, since this error
component is used to bring about the correction. This component is called the error signal.
The error, or deviation from the desired signal
193
is passed through the feedback loop to cause
an adjustment to reduce the value of the error
signal. Care must be taken in the design of
the feedback loop to reduce over -control tendencies wherein the correction signal would
carry the sytem past the point of correct operation. Under certain circumstances the new
error signal would cause the feedback control
to overcorrect in the opposite direction, resulting in hunting or oscillation of the closed
loop system about the correct operating point.
Negative feedback control would tend to
dampout spurious system oscillation if it were
not for the time lag or phase shift in the system. If the overall phase shift is equal to one half cycle of the operating frequency of the
system the feedback will maintain a steady
state of oscillation when the circuit gain is
sufficiently high, as shown in figure 21. In
order to prevent oscillation, the gain figure of
the feedback loop must be less than unity when
the phase shift of the system reaches 180 degrees. In an ideal control system the gain of
the loop would be constant throughout the
operating range of the device, and would drop
rapidly outside the range to reduce the bandwidth of the control system to a minimum.
The time lag in a closed loop system may
reduced by using electronic circuits in place
of mechanical devices, or by the use of special
circuit elements having a phase -lead characteristic. Such devices make use of the properties of a capacitor, wherein the current leads
the voltage applied to it.
be
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CHAPTER ELEVEN
Electronic Computers
Mechanical computing machines were first
produced in the seventeenth century in Europe although the simple Chinese abacus (a
digital computer) had been in use for centuries. Until the last decade only simple mechanical computers (such as adding and bookkeeping machines) were in general use.
The transformation and transmission of the
volume of information required by modern
CIIP
011,
technology requires that machines assume many
of the information processing systems formerly done by the human mind. Computing machines can perform routine operations more
quickly and more accurately than a human being, processing mathematical and logistical
data on a production line basis. The computer,
however, cannot create, but can only follow
instructions. If the instructions are in error,
0111
THE IBM
COMPUTER
AND
"MEMORY"
The
puter
"704" Comis used
32,000
with
"word"
storage
memory
unit for research
Heart
programs.
of this auxiliary
unit are small,
doughnut- shaped
iron ferrites which
store information
by means of magnetism. The unit
is the first of Ms
kind to be installed with I B M's
704 computer.
Components of the
system seen in the
foreground are
(left) card punch
and (right) cord
.j
reader. In the
center of the picture is the 704's
processing unit.
www.americanradiohistory.com
Digital Computers
195
NORTN SNORE
FARN!R
CORN
N!N
^FOX
NOTE: ALL BUTTONS NAVE
ONE NORMALLY OPEN CONTACT AND
ONE NORMALLY CLOSED CONTACT.
Figure 2
A SEQUENCE COMPUTER.
Three correct buttons will sound the buzzer.
SOUTN SNORE
Figure 1
SIMPLE PUZZLES IN LOGIC MAY BE
SOLVED BY ELECTRIC COMPUTER.
THE "FARMER AND RIVER"
COMPUTER IS SHOWN HERE.
the computer will produce a wrong answer.
Computers may be divided into two classes:
the digital and the analog. The digital computer counts, and its accuracy is limited only by
the number of significant figures provided for
in the instrument. The analog computer
measures, and its accuracy is limited by the
percentage errors of the devices used, multiplied by the range of the variables they represent.
11 -1
Digital Computers
The digital computer operates in discrete
steps. In general, the mathematical operations
are performed by combinations of additions.
Thus multiplication is performed by repeated
additions, and integration is performed by
summation. The digital computer may be
thought of as an "on -off" device operating
from signals that either exist or do not exist.
The common adding machine is a simple computer of this type. The "on -off" or "yes-no"
type of situation is well suited to switches, electrical relays, or to electronic tubes.
A simple electrical digital computer may be
used to solve the old "farmer and river" problem. The farmer must transport a hen, a bushel
of corn, and a fox across a river in a small
boat capable of carrying the farmer plus one
other article. If the farmer takes the fox in
the boat with him, the hen will eat the corn.
On the other hand, if he takes the corn, the
fox will eat the hen. The circuit for a simple
computer to solve this problem is shown in
figure 1. When the switches are moved from
"south shore" to "north shore" in the proper
sequence the warning buzzer will not sound.
An error of choice will sound the buzzer.
A second simple "digital computer" is shown
is figure 2. The problem is to find the three
proper push buttons that will sound the buzzer.
The nine buttons are mounted on a board so
that the wiring cannot be seen.
Each switch of these simple computers executes an "on -off" action. When applied to a
logical problem "yes-no" may be substituted
for this term. The computer thus can act out
a logical concept concerned with a simple
choice. An electronic switch (tube) may be
substituted for the mechanical switch to increase the speed of the computer. The early
computers, such as the ENIAC (Electronic Numerical Integrator and Calculator) employed
over 18,000 tubes for memory and registering
circuits capable of "remembering" a 10 -digit
number.
11 -2
Binary Notation
To simply and reduce the cost of the digital
computer it was necessary to modify the system
of operation so that fewer tubes were used per
bit of information. The ENIAC -type computer
requires 50 tubes to register a 5 -digit number.
O
0
0
0
O
O
O
O
O
O
O
O
O
O
O
O
O
O
O
O
O
O
O
{S.
O
O
O
O
O
O
O
O
O
O
O
O
:;oFigure
0
O
0
0
O
O
O
O
3
BINARY NOTATION MAY BE USED
FOR DIGITAL DISPLAY. BINARY
BOARD ABOVE INDICATES "73092."
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THE RADIO
Electronic Computers
196
O
O
-`
:O"--
BINARY NOTATION
0
o
TUBE(S)
DIGIT
I
1
a
a
3
2+1
4
4
5
+1
7
+2+1
s
4
9
!+1
10
6+2
1/
5+2+1
12
+
+4+1
14
+4+2
15
+4+2+1
1
1
2
1,0
3
1,1
s
1,0.0
1,0,1
1.1.0
6
-
4+2
12
DECIMAL NOTATION
,
7
1.1.1
0
1,0,0,0
1,0,0,1
1,0,1,0
0
10
Figure 5
BINARY NOTATION SYSTEM
REQUIRES ONLY TWO NUMBERS,
"0" AND "1."
Figure 4
BINARY DECIMAL NOTATION. ONLY
FOUR TUBES ARE REQUIRED TO
REPRESENT DIGITS FROM
TO 15.
THE DIGIT "12" IS INDICATED
1
ABOVE.
The tubes (or their indicator lamps) can be
arranged in five columns of 10 tubes each.
From right to left the columns represent units,
tens, hundreds, thousands, etc. The bottom tube
in each column represents "zero," the second
tube represents "one," the third tube "two,"
and so on. Only one tube in each column is
excited at any given instant. If the number
73092 is to be displayed, tube number seven
in the fifth column is excited, tube number
three in the fourth column, tube number zero
in the third column, etc. as shown in figure 3.
A simpler system employs the binary decimal notation, wherein any number from one
to fifteen can be represented by four tubes.
Each of the four tubes has a numerical value
that is associated with its position in the tube
group. More than one tube of the group may
be excited at once, as illustrated in figure 4.
The values assigned to the tubes in this particular group are 1, 2, 4, and 8. Additional
tubes may be added to the group, doubling the
notation of the rube thus: 1, 2, 4, 8, 16, 32,
64, 128, 356, etc. Any numerical value lower
than the highest group number can be displayed by the correct tube combination.
A third system employs the binary notation
which makes use of a bit (binary digit) representing a single morsel of information. The
binary system has been known for over forty
centuries, and was considered a mystical revelation for ages since it employed only two sym-
bols for all numbers. Computer service usually employs "zero" and "one" as these symbols.
Decimal notation and binary notation for common numbers are shown in figure 5. The
binary notation represents 4 -digit numbers
(thousands) with .ten bits, and 7 -digit numbers (millions) with 20 bits. Only one electron tube is required to display an information
bit. The savings in components and primary
power drain of a binary -type computer over
the older ENIAC -type computer is obvious.
Figure 6 illustrates a computer board showing the binary indications from one to ten.
Digital
Computer
The digital computer is em-
ployed in a "yes -no" situation. It may be used for
routine calculations that would ordinarily require enormous man -hours of time, such as
checking stress estimates in aircraft design, or
military logistics, and problems involving the
manipulation of large masses of figures.
Uses
DECIMAL NOTATION
COMPUTER NOTATION
o
O
1
o
o
a
3
0
5
0
o
4
O
O
o
0
0
0O
O
4
7
.
o
0
0
+
10
0
=
Figure
o
OFF
6
BINARY NOTATION AS REPRESENTED
ON COMPUTER BOARD FOR NUMBERS
FROM 1 TO 10.
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HANDBOOK
Analog Computers
eour=e,+e2
197
+150V.
e Our
R,
e,
eour-
R,
R, +R2
e2 R2
Ri +R2
/
R3
R, R2
\[R,+ R234-R3J
Figure 7
OF TWO VOLTAGES
BY ELECTRICAL MEANS.
SUMMATION
Analog Computers
11 -3
The analog computer represents the use of
one physical system as a model for a second
system that is usually more difficult to construct or to measure, and that obeys the equations of the same form. The term analog implies similarity of relations or properties between the two systems. The common slide-rule
is a mechanical analog computer. The speedometer in an automobile is a differential analog
computer, displaying information proportional
to the rate of change of speed of the vehicle.
The electronic analog computer employs circuits containing resistance, capacitance, and inductance arranged to behave in accordance with
analogous equations. Variables are represented
by d -c voltages which may vary with time.
Figure 8
OF TWO VOLTAGES
BY ELECTRONIC MEANS.
SUMMATION
Thus complicated problems can be solved by
d -c amplifiers and potentiometer controls in
electronic circuits performing mathematical
functions.
If a linear network is energized by two voltage sources
the voltages may be summed
as shown in figure 7. Subtraction of quantities
may be accomplished by using negative and
positive voltages. A -c voltages may be employed for certain additive circuits, and more
Addition and
Subtraction
THE
HEATHKIT
ELECTRONIC
ANALOG
DIGITAL
COMPUTER
"electronic
slide rule" simuThis
lates equations or
physical problems
electronically, sub-
stituting one phys-
ical system as a
model for a sec-
system that
usually more
or costly
to construct or
measure, and that
obeys equations of
the some form.
ond
is
difficult
www.americanradiohistory.com
eour
THE RADIO
Electronic Computers
198
R2 .
tt
=
c
e
eour
Figure 9
ELECTRONIC MULTIPLICATION
MAY BE ACCOMPLISHED BY
CALIBRATED POTENTIOMETERS,
WHEN OUTPUT VOLTAGE IS
PROPORTIONAL TO THE INPUT
VOLTAGE MULTIPLIED BY A
CONSTANT (R R1).
complex circuits employ vacuum tubes, as in
figure 8. Synchronous transformers may be
used to add expressions of angular rotation,
and circuits have been developed for adding
time delays, or pulse counting.
Multiplication
Electronic multiplication and
division may be accomplished
with the use of potentiometers where the output voltage is proportional to
the input voltage multiplied by a constant
which may be altered by changing the physical
arrangement of the potentiometer (figure 9).
Variable autotransformers may also be used to
perform multiplication.
A simple bridge may be used to obtain an
output that is the product of two inputs divided
by a third input, as shown in figure 10.
and Division
Differentiation
The time derivative of a
voltage can be expressed as
a charge on a capacitor by:
de
(1)
dt
and is shown in figure 11A. The charging current is converted into a voltage by the use
of a resistor, R. If the input to the RC circuit
is charging at a uniform rate so that the current through C and R is constant, the output
voltage e is:
i
OUTPUT'\,
=C
e
our
Figure 11
ELECTRONIC DIFFERENTIATION
The time derivative of a voltage can be
expressed as a charge on a capacitor (A).
Operational amplifier (B) employs feed
back principle for short differentiation
time.
RCd
(2)
For highest accuracy, a small RC product
should be used, permitting the maximum possible differentiation time. The output of the
differentiator may be amplified to any suitable
level.
A more accurate differentiating device
makes use of an operational amplifier. This
unit is a high gain, negative feedback d -c
amplifier (discussed in section 11 -4) with the
resistance portion of the RC product appearing
in the feedback loop of the amplifier (figure
11B). A shorter differentiation time may be
employed if the junction point between R and
C could be held at a constant potential. The
feedback amplifier shown inverts the output
signal and applies it to the RC network, hold-
ing the junction potential constant.
Integration
Integration is a process of accumulation, or summation, and
requires a device capable of storing physical
quantities. A capacitor will store an electrical
charge and will give the time integral of a
current in respect to a voltage:
eo= 1
idt
(3)
In most computers, the input signal is in
the form of a voltage, and the input charging
current of the capacitor must be taken through
a series resistance as in figure 12. If the integrating time is short the charging current is
approximately proportional to the input voltage. The charging current may be made a
true measure of the input voltage by the use
RI x R3
R2
Figure 10
ELECTRONIC MULTIPLICATION BY
BRIDGE CIRCUIT PROVIDES
OUTPUT THAT IS PRODUCT OF
TWO INPUTS DIVIDED BY A
THIRD INPUT.
Figure 12
SIMPLE
INTEGRATION
CIRCUIT
Making use of
charging current
of capacitor.
www.americanradiohistory.com
e
O
T
c
eouT
HANDBOOK
Operational Amplifier
-
The Operational
-4
11
199
Amplifier
Mathematical operations are performed by
using a high gain d -c amplifier, termed an
operational amplifier. The symbol of this unit
is a triangle, with the apex pointing in the direction of operation ( figure 15). The gain of
such an amplifier is -A, so:
+
Figure 3
"MILLER FEEDBACK" INTEGRATOR
SUITABLE FOR COMPUTER USE.
eaur
eo=-Aeg,oreg=
Figure 14
R -L NETWORK USED FOR
INTEGRATION PURPOSES.
-e
(4)
A
If -A approaches infinity, e, will be approximately zero. In practice this condition is
realized by using amplifiers having open loop
gains of 30,000 to 60,000. If ea is set at 100
volts, e, will be of the order of a few millivolts. Thus, considering eg equal to zero:
o
Figure 15
OPERATIONAL AMPLIFIER ( -A)
Mathematical operations may be performed
by any operational amplifier, usually a
stable, high-gain d -c amplifier, such as
RI
-
Rf
,oreo-
Rr
RI
(5)
eI
which may be written:
shown in Figure 16.
e0
of an operational amplifier wherein the capacitance portion of the RC product appears
in the feedback loop of the amplifier, holding
the junction point between R and C at a constant potential. A simple integrator is shown
in figure 13 employing the Miller feedback
principle. Integration is also possible with an
RL network (figure 14).
_
-
Keg, where K
=
R
I
(6 )
This amounts to multiplication by a constant
coefficient, since RI and Rr may be fixed in
value. The circuit of a typical operational amplifier is shown in figure 16.
Amplifier
Operation
Two voltages may be added by
the amplifier, as shown in figure
17. Keeping in mind that eg is
u (Rv)
e,
BIAS
-4í0V
GAIN=
-ze0 V
-A
Figure 16
HIGH GAIN OPERATIONAL AMPLIFIER, SUCH AS USED IN HEATH COMPUTER.
200
THE RADIO
Electronic Computers
RF
o
e
ea
D
o
eo
Figure 17
TWO VOLTAGES MAY BE
ADDED BY SUMMATION
AMPLIFIER.
F
essentially at zero (ground) potential:
eo= eo
or,
where
Re
=
R.
K.
K=
e
Re
+
e,+K:e
(8)
and K: =
B
(7)
e2
R3
R`
As long as e., does not exceed the input
range of the amplifier, any number of inputs
may be used:
eo=
-
Re
Rl
ei
Rf
ei +
+ Á=
- - -
+
RI
Ro
en
11
By combining the above operations in various ways, problems of many kinds may be
solved .For example, consider the mass- springdamper assembly shown in figure 19. The mass
M is connected to the spring which has an
elastic constant K. The viscous damping constant is C. The vertical displacement is y. The
sum of. the forces acting on mass M is:
f
or
eo
= -Rr+
(11)
Ro
Integration is performed by replacing the
feedback resistor Re with a capacitor Ce, as
shown in figure 18. For this circuit (with ea
approximately zero):
t
but g=Cr
en, so
dc
ei
dt
Ri
d=
Ce
do
(12)
and
R
= Cf
1
and
o
e
o
R 1Cr eidt
deo
eo
ei dt
Ri Cr
(t) =M
d$
+
C
d
+Ky
(14)
(15)
.
M
dt'
-C
dt
-Ky +f (t)
(17)
If, in the analog circuit, there is a voltage
-dt
equal to M
it can be converted to
dby passing it through an integrator circuit having an RC time constant equal to M. This resulting voltage can be passed through a second
integrator stage with unit time constant which
will haue an output volage equal to y. The
voltages representing y,
Figure 18
INTEGRATION
RI
AN
oeo
o
Performed by
Summation Amplifier
by replacing feed
back resistor with a
capacitor.
(16)
where f (t) is the applied force, or forcing
function.
The first step is to set up the analog computer circuit so as to obtain an output voltage
proportional to y for a given input voltage
proportional to f (t) Equation (16) may be
rewritten in the form:
deo
(13)
Thus:
Solving Analog
Problems
-5
(10)
en
=eft/
Figure 19
"MASS- SPRINGDAMPER" PROBLEM
MAY BE SOLVED BY
ELECTRICAL ANALOGY
WITH SIMPLE
COMPUTER.
then be summed to give
(t)
can
- Ky +
f (t)
and f
-C
do
which is the right hand side of equation (17) ,
r
and therefore equal to M
www.americanradiohistory.com
dt
.
Connecting the
HANDBOOK
Analog Problems
201
SET VOLTAGE TO Y =YO As
TIME
(INITIAL
t'0
M
dZ
-2.dr
-Cdt -Kr+f(t
dt
dt 1
DISPLACEMENT)
DISPLAY
OSCILLOSCOPE
TO
Cdt - fit)
F(t)
Figure 20
ANALOG SOLUTION FOR "MASS- SPRINGDAMPER" PROBLEM OF FIGURE 19.
output of the summing amplifier (A3) to the
input of the first as shown in figure 20 satisfies the equation.
To obtain a solution to the problem, the
initial displacement and velocity must be specified. This is done by charging the integrating
capacitors to the proper voltages. Three operational amplifiers and a summing amplifier
are required.
A second problem that may be solved by
the analog computer is the example of a freely
falling body. Disregarding air resistance, the
body will fall (due to the action of gravity)
with a constant acceleration. The equation
describing this action is:
F
=mg =m
(18)
dt'
Integration of equation (18) will give the
velocity, or
,
dt
and integration a second time
will give displacement, or y. The block diagram of a suitable computer for this problem
is shown in figure 21.
If a voltage proportional to g and hence to
d'
dY
is
introduced into the first amplifier, the
-
output of that unit will he
-H
,
or the
--R2
Ri
e,=
dt
dZr
dtZ
r
-dr
dt
eo
G
Figure 21
ANALOG COMPUTER FOR
"FREELY FALLING BODY"
PROBLEM.
velocity. That, in turn, will become y, or distance, at the output of the second amplifier.
Before the problem can be solved on the
computer it is necessary to determine the time
of solution desirable and the output amplitude
of the solution. The time of solution is determined by the RC constant of the integrating
amplifiers. If RC is set at unity, computer
time is equal to real time. The computer time
desirable is determined by the method of readout. When using an oscilloscope for read -out,
a short solution time is desirable. For a recorder, longer solution time is better.
Suppose, for example, in the problem of the
falling body, the distance of fall in 2.5 seconds
is desired. Using an RC constant of 1 would
give a solution time of 2.5 seconds. This would
be acceptable for a recorder but is slow for an
oscilloscope. A convenient time of solution for
the 'scope would be 25 milliseconds. This is
1/100 of the real time, so an RC constant of
.01 is needed. This can be obtained with C
equal to 0.1 pfd, and R equal to 100,000
ohms.
It is now necessary to choose an input voltage which will not overdrive the amplifiers.
The value of g is known to be approximately
32 ft /sec. /sec. A check indicates that if we
set g equal to 32 volts, the voltage representing the answer will exceed 100 volts. Since
the linear response of the amplifier is only
100 volts, this is undesirable. An input of 16
volts, however, should permit satisfactory operation of the amplifiers. Output voltages near
zero should also be avoided. In general, output
voltage should be about 50 volts or so, with
amplifier gains of 20 to 60 being preferable.
Thus, for this particular problem the time scale factor and amplitude -scale factor have
www.americanradiohistory.com
202
THE RADIO
Electronic Computers
A'
A RELAY
RELAY
100 K
d2r
16 V.
dt2
dr
eo= Y
Figure 22
ANALOG SOLUTION FOR
"FALLING BODY PROBLEM"
OF FIGURE 21.
Figure 24
LIMITING CIRCUIT TO SIMULATE
NON -LINEAR FUNCTIONS SUCH AS
TIME IN SECONDS
4
eo
e,
tr
dt
(c
ENCOUNTERED IN HYSTERESIS,
BACKLASH, AND FRICTION
PROBLEMS.
s)
6
16
eo
32
VOLTS
DISTANCE
N
11 -6
FEET
-24
46
1
-32
Problems are frequently encountered in
which non -linear functions must be simulated.
Non -linear potentiometers may be used to supply an unusual voltage source, or diodes may
be used as limiters in those problems in which
a function is defined differently for different
regions of the independent variable. Such a
function might be defined as follows:
64
-40
-80
)
_'98
-46
36
1'00)
112
64
28
72
,44
Non -linear Functions
Figure 23
READ -OUT SOLUTION OF "FREELY
FALLING BODY" PROBLEM.
e..
e..
e..
been chosen. The problem now looks like
figure 22.
To solve the problem, relays A and A' are
opened. The solution should now appear on
the oscilloscope as shown in figure 23. The
solution of the problem leaves the integrating
capacitors charged. It is necessary to remove
this charge before the problem can be rerun.
This is done by closing relays A and A'.
--
- K:
=
K1, e,
= et, KT,, e_,
= K2, el ,, K:
KI
(19)
(20)
(21)
where K, and K: are constants.
Various limiting circuits can be used, one of
which is shown in figure 24. This is a series
limiter circuit which is simple and does not
require special components. Commonly encountered problems requiring these or similar
limiting techniques include hysteresis, backlash, and certain types of friction.
C
NOTE: REPLACE
C WITN A I MEC. RESISTOR
FOR FUNCTION SETUP
MEG
X
OUTPUT
Y
OUTPUT
RAMP - FUNCTION
GENERATOR
620
K
P
SLOPE CONTROL
E
1
2 6AL5
BREAK
CONTROL
VOLTAGE
620K
t
SIGN CHANGING
AMPLIFIER
SUMMING
AMPLIFIER
Figure 25
SIMPLIFIED
DIAGRAM OF FUNCTIONAL GENERATOR TO APPROXIMATE NON- LINEAR
FUNCTIONS.
HANDBOOK
XI
o
Xz
Non -Linear Functions
7X3
Figure 26
TYPICAL NON -LINEAR FUNCTION
WHICH MAY BE SET UP WITH
FUNCTION GENERATOR.
The Function
Generator
function generator may be
used to approximate almost
any non -linear function. This
is done by use of straight line segments which
are combined to approximate curves such as
are found in trigonometric functions as well
as in stepped functions. In a typical generator
ten line segments are used, five in the plus -x
direction, and five in the minus -x direction.
Five 6ÁL5 double diodes are used. Each line
segment is generated by a modified bridge
circuit (figure 25). A ramp function or voltage is fed into one arm of the bridge while
the opposite arm is connected to a biased diode.
The other two arms of the bridge combine to
form the output. The voltage appearing across
one of these arms is fed through a sign- changing amplifier and then summed with the voltage appearing at the opposite arm. If the arm
of potentiometer P (the slope control) is set
in the center, the bridge will be balanced and
A
Figure 27
ELECTRONIC PACKAGE IN
DIGITAL COMPUTER.
Stylized diagram of tube package. Lines carrying negative pulses are marked by a small
circle at each end. Gates are indicated by
a semi -circle with "pins" for each input.
the output of the summing amplifier will be
zero. If, on the other hand, potentiometer P
is adjusted one way or the other from center,
the bridge will be unbalanced and the summing amplifier output will vary linearly with
respect to the input in either a positive or negative y direction, depending upon which side
of center potentiometer P is set.
The break voltage, or value of x at which a
straight line segment will begin is set by
biasing the diode to the particular voltage level
or value of x desired. The ramp function generator has either a positive or negative input
which because of the 180 degree phase shift
in the amplifier, gives a minus- or plus -x output respectively. A typical function such as
shown in figure 26 may be set up with the
function generator. The initial condition volt-
IBM's new "608," the
first completely transistorized calculator
for commercial applications, operates
without the use of a
single vacuum tube.
Transistors - -tiny germanium devices that
perform many of the
functions of conventional vacuum tubes
-make
possible 50%
reduction in computer -unit size and a
in
reduction
90%
requirements
power
over a comparable
IBM tube -model machine.
They
203
are
mounted, along with
related circuitry, on
banks of printed wiring panels in the
608.
The machine's interstorage,
or "memnal
ory," is made up of
magnetic cores -minute, doughnut -shaped objects that can
"remember" information indefinitely, and
recall it for use in
calculations in a few
millionths of a
second.
www.americanradiohistory.com
204
THE RADIO
Electronic Computers
age is set to the value of X. The break -voltage
control is increased until the output of the
summing amplifier increases abruptly, indicating the diode is conducting. The input
voltage from the initial condition power supply
is set to the value X: The slope control (P)
is now set to value Y2. A second function generator may be used to set points X.1 and X.,
using the break -voltage control and the initial
condition voltage adjustments. Points XS and
X. are finally set with a third generator. The
x- output of the function generator system may
be read on an oscilloscope, using the x- output
of the ramp- function generator amplifier as
the horizontal sweep for the oscilloscope.
Digital Circuitry
11 -7
Digital circuits dealing with "and," "or,"
and "not" situations may be excited by electrical pulses representing these logical operations.
Sorting and amplifying the pulses can be accomplished by the use of electronic packages,
such as shown in figure 27. Logical operations
may be accomplished by diode -resistor gates
operating into an amplifier stage. Negative and
positive output pulses from the amplifier are
obtained through diode output gates. The driving pulses may be obtained from a standard
oscillator, operating at or near 1 mc.
A circuit of a single digital package is shown
in figure 28. Other configurations, such as a
"flip-flop" may be used. Many such packages
"AND
can be connected in series to form operational
circuits. The input "and" and "or" gates are
biased to conduction by external voltages. The
"and" diode gate transmits a pulse only when
all the input terminals are pulsed positively,
and the "or" diode gate transmits a positive
pulse applied to any one of its input terminals.
The input pulses pass through the gates and
drive the amplifier stage, which delivers an
amplified pulse to the positive and negative
output gates, and to accompanying memory
circuits.
Memory Circuits
A memory circuit consists
of some sort of delay line
which is capable of holding an information
pulse for a period of time. The amount of
delay is proportional to the frequency of the
input signal. A "long" transmission line may
be used as a delay line with the signal being
removed from the "far" end of the line after
being delayed an interval equal to the time
of transmission along the line. Lines of this
type are constructed in the manner of a
coaxial cable, except that the inner conductor
is a long, thin coil of wire. Other memory
circuits make use of magnetostrictive or piezoelectric effects to retard the pulse. Information
may also be stored in electrostatic storage
tubes, upon magnetic recording tape, and in
ferro- magnetic cores capable of holding 10,000
bits of information.
GATES
INPUT SI
LIMITING
INPUT
DIODES
2
"OR"
GATE
INPUTSS
INPUT
*4
INPUT
*5
INPUTS
6
INPUT*
7
CLAMPING
DIODES
6AN5
POSITIVE PULSE
0
NEGATIVE PULSE
INPUT
Figure 28
TYPICAL DIGITAL PACKAGE SHOWING INPUT AND OUTPUT DIODE GATES AND
PULSE AMPLIFIER.
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CHAPTER TWELVE
Radio Receiver Fundamentals
A conventional reproducing device such as
loudspeaker or a pair of earphones is incapable of receiving directly the intelligence
carried by the carrier wave of a radio transmitting station. It is necessary that an additional device, called a radio receiver, be
placed between the receiving antenna and the
loudspeaker or headphones.
Radio receivers vary widely in their complexity and basic design, depending upon the
intended application and upon economic factors. A simple radio receiver for reception of
radiotelephone signals can consist of an earphone, a silicon or germanium crystal as a
carrier rectifier or demodulator, and a length
of wire as an antenna. However, such a receiver is highly insensitive, and offers no
significant discrimination between two signals in the same portion of the spectrum.
On the other hand, a dual -diversity receiver
designed for single -sideband reception and
employing double or triple detection might
occupy several relay racks and would cost
many thousands of dollars. Ilowever, conventional communications receivers are intermediate in complexity and performance between
the two extremes. This chapter is devoted to
the principles underlying the operation of
such conventional communications receivers.
12 -1
a
Detection or
Demodulation
A detector or demodulator is a device for
removing the modulation (demodulating) or
detecting the intelligence carried by an incoming radio wave.
Figure 1 illustrates an elementary form of radiotelephony receiver employing a
diode detector. Energy from a passing radio
wave will induce a voltage in the antenna and
cause a radio- frequency current to flow from
antenna to ground through coil Lt. The alternating magnetic field set up around L, links
with the turns of L2 and causes an r -f current
to flow through the parallel -tuned circuit,
1..2-C1. %hen variable capacitor C, is adjusted
so that the tuned circuit is resonant at the
frequency of the applied signal, the r-f voltage
is maximum. This r -f voltage is applied to the
diode detector where it is rectified into a varying direct current and passed through the earphones. The variations in this current correspond to the voice modulation placed on the
signal at the transmitter. As the earphone
diaphragms vibrate back and forth in accordRadiotelephony
Demodulation
205
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206
Radio Receiver Fundamentals
THE
RADIO
TRIODE
lquirlli
'Ili
O
AUDIO OUTPUT
-
GROUND
L+
L2
I-
e
+
PLATE- TICKLER REGENERATION WITH "THROTTLE'
CONDENSER REGENERATION CONTROL.
Figure
ELEMENTARY FORM OF RECEIVER
This is the basis of the "crystal set" type of
1
PENTODE
ceiver, although a vacuum diode may be used in
place of the crystal diode. The tank circuit L2 -C1
is tuned to the frequency it is desired to receive.
The bypass capacitor across the phones should
have a low reactance to the carrier frequency being received, but a high reactance to the modulation on the received signal.
ance with the pulsating current they audibly
reproduce the modulation which was placed
upon the carrier wave.
The operation of the detector circuit is
shown graphically above the detector circuit
in figure 1. The modulated carrier is shown
at A, as it is applied to the antenna. B represents the same carrier, increased in amplitude,
as it appears across the tuned circuit. In C
the varying d -c output from the detector is
seen.
Radiotelegraphy
Reception
AUDIO OUTPUT
re-
+e
-e
CATHODE-TAP REGENERATION WITH SCREEN VOLTAGE
REGENERATION CONTROL.
Figure 2
REGENERATIVE DETECTOR CIRCUITS
Regenerative detectors are seldom used at the present time due to their poor selectivity. However,
they do illustrate the simplest type of receiver
which may be used either for radiophone or radiotelegraph reception.
Since a c -w telegraphy sig-
nal consists of an unmodulated carrier which is interrupted to form dots and dashes, it is apparent
that such a signal would not be made audible
by detection alone. While the keying is a form
of modulation, it is composed of such low frequency components that the keying envelope
itself is below the audible range for hand keying speeds. Some means must be provided
whereby an audible tone is heard while the
unmodulated carrier is being received, the tone
stopping immediately when the carrier is in-
terrupted.
The most simple means of accomplishing
this is to feed a locally generated carrier of
a slightly different frequency into the same
detector, so that the incoming signal will mix
with it to form an audible beat note. The difference frequency, or heterodyne as the beat
note is known, will of course stop and start
in accordance with the incoming c -w radiotelegraph signal, because the audible heterodyne can exist only when both the incoming
and the locally generated carriers are present.
The Autodyne
Detector
The local signal which is used
to beat with the desired c -w
signal in the detector may be
supplied by a separate low-power oscillator
in the receiver itself, or the detector may be
made to self -oscillate, and thus serve the
dual purpose of detector and oscillator. A detector which self -oscillates to provide a beat
note is known as an autodyne detector, and
the process of obtaining feedback between the
detector plate and grid is called regeneration.
An autodyne detector is most sensitive when
it is barely oscillating, and for this reason
a regeneration control is always included in
the circuit to adjust the feedback to the proper
amount. The regeneration control may be either
a variable capacitor or a variable resistor,
as shown in figure 2.
With the detector regenerative but not oscillating, it is also quite sensitive. When the
circuit is adjusted to operate in this manner,
modulated signals may be received with considerably greater strength than with a non regenerative detector.
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Superregenerative Detectors
HANDBOOK
12 -2
Superregenerative
Receivers
ultra -high frequencies, when it is desired to keep weight and cost at a minimum,
a special form of the regenerative receiver
known as the superregenerator is often used
for radiotelephony reception. The superregenerator is essentially a regenerative receiver
with a means provided to throw the detector
rapidly in and out of oscillation. The frequency
at which the detector is made to go in and out
of oscillation varies with the frequency to be
received, but is usually between 20,000 and
500,000 times a second. This superregenerative action considerably increases the sensitivity of the oscillating detector so that the
usual "background hiss" is greatly amplified
when no signal is being received. This hiss
diminishes in proportion to the strength of the
received signal, loud signals eliminating the
hiss entirely.
TO
At
There are two systems in common
use for causing the detector to break
in and out of oscillation rapidly. In
one, a separate interruption -frequency oscillator is arranged so as to vary the voltage rapidly on one of the detector tube elements (usually the plate, sometimes the screen) at the high
rate necessary. The interruption- frequency
oscillator commonly uses a conventional tickler- feedback circuit with coils appropriate for
its operating frequency.
The second, and simplest, type of super regenerative detector circuit is arranged so
as to produce its own interruption frequency
oscillation, without the aid of a separate tube.
The detector tube damps (or "quenches ") itself
out of signal- frequency oscillation at a high
rate by virtue of the use of a high value of
grid leak and proper size plate- blocking and
grid capacitors, in conjunction with an excess
of feedback. In this type of "self- quenched"
detector, the grid leak is quite often returned
to the positive side of the power supply (through
the coil) rather than to the cathode. A representative self-quenched superregenerative detector circuit is shown in figure 3.
Except where it is impossible to secure
sufficient regenerative feedback to permit
superregeneration, the self-quenching circuit
is to be preferred; it is simpler, is self- adjusting as regards quenching amplitude, and can
have good quenching wave form. To obtain
as good results with a separately quenched
superregenerator, very careful design is required. However, separately quenched circuits
are useful when it is possible to make a certain tube oscillate on a very high frequency
but it is impossible to obtain enough regeneration for self-quenching action.
Quench
Methods
2J7
AUDIO
AMPLIFIER
Figure 3
SUPERREGENERATIVE DETECTOR CIRCUIT
A sell -quenched superregenerative detector such
as illustrated above is capable of giving good
sensitivity in the v-h -f range. However, the circuit
has the disadvantage that its selectivity is relatively poor. Also, such o circuit should be preceded by an r -I stage to suppress the radiation of
o signal by the oscillating detector.
The optimum quenching frequency is a function of the signal frequency. As the operating
frequency goes up, so does the optimum quenching frequency. hen the quench frequency is
too low, maximum sensitivity is not obtained.
When it is too high, both sensitivity and selectivity suffer. In fact, the optimum quench frequency for an operating frequency below 15 Mc.
is in the audible range. This makes the superregenerator impracticable for use on the lower
frequencies.
The high background noise or hiss which
is heard on a properly designed superregenerator when no signal is being received is not
the quench frequency component; it is tube
and tuned circuit fluctuation noise, indicating
that the receiver is extremely sensitive.
A moderately strong signal will cause the
background noise to disappear completely,
because the superregenerator has an inherent
and instantaneous automatic volume control
characteristic. This same a -v-c characteristic
makes the receiver comparatively insensitive
to impulse noise such as ignition pulses -a
highly desirable feature. This characteristic
also results in appreciable distortion of a received radiotelephone signal, but not enough
to affect the intelligibility.
The selectivity of a superregenerator is
rather poor as compared to a superheterodyne,
but is surprisingly good for so simple a receiver when figured on a percentage basis
rather than absolute kc. bandwidth.
FM Reception
A.
superregenerative receiver
will receive frequency modulated signals with results comparing favorably
with amplitude modulation if the frequency
swing of the FM transmitter is sufficiently
high. For such reception, the receiver is detuned slightly to either side of resonance.
www.americanradiohistory.com
208
Radio
Receiver Fundamentals
THE
RADI
O
AUDIO
OUTPUT
22 MC
OUTPUT
IINTERMED.
RF
AMPLIFIER
I
"SECOND*
REOUENCV
AMPLIFIER
I_
AUDIO
'AMPLIFIER
DETECTOR
1
I
-
0+100V
FREONCY
IOSCILLATORI
(FOR C.W.)
I
AUOIO
Figure 5
ESSENTIAL UNITS OF A
SUPERHETERODYNE RECEIVER
The basic portions of the receiver are shown
in solid blocks. Practicable receivers employ the dotted blocks and also usually include such additional circuits os a noise
limiter,
Figure 4
THE FREMODYNE SUPERREGENERATIVE
SUPERHETERODYNE DETECTOR FOR
FREQUENCY MODULATED SIGNALS
Superregenerative receivers radiate a strong,
broad, and rough signal. For this reason, it is
necessary in most applications to employ a
radio frequency amplifier stage ahead of the
detector, with thorough shielding throughout
the receiver.
The Fremodyne
Detector
The Hazel tin e- Fremodyne
superregenerative circuit is
expressly designed for re-
ception of FM signals. This versatile circuit
combines the action of the superregenerative
receiver with the superhetrodyne, converting
FM signals directly into audio signals in one
double triode tube (figure 4). One section of
the triode serves as a superregenerative mixer,
producing an i -f of 22 Mc., an i -f amplifier, and
a FM detector. The detector action is accomplished by slope detection tuning on the side
of the i -f selectivity curve.
This circuit greatly reduces the radiated
signal, characteristic of the superregenerative
detector, yet provides many of the desirable
features of the superregenerator. The pass band of the Fremodyne detector is about
400 kc.
12 -3
Superheterodyne
Receivers
Because of its superiority and nearly universal use in all fields of radio reception, the
circuit, and a crystal
in the i -f amplifier.
on a -v -c
filter
theory of operation of the superheterodyne
should be familiar to every radio student and
experimenter. The following discussion concerns superheterodynes for amplitude- modulation reception. It is, however, applicable in
part to receivers for frequency modulation.
Principle of
In the superheterodyne, the incoming signal is applied to a
mixer consisting of a non -linear
impedance such as a vacuum tube or a diode.
The signal is mixed with a steady signal generated locally in an oscillator stage, with the
result that a signal bearing all the modulation
applied to the original signal but of a frequency equal to the difference between the
local oscillator and incoming signal frequencies appears in the mixer output circuit. The
output from the mixer stage is fed into a fixed tuned intermediate -frequency amplifier, where
it is amplified and detected in the usual manner, and passed on to the audio amplifier. Figure 5 shows a block diagram of the fundamental superheterodyne arrangement. The basic
components are shown in heavy lines, the
simplest superheterodyne consisting simply
of these three units. However, a good communications receiver will comprise all of the
elements shown, both heavy and dotted blocks.
Operation
Superheterodyne
Advantages
The advantages of super heterodyne reception are
directly attributable to the
use of the fixed -tuned intermediate -frequency
(i -f) amplifier. Since all signals are converted
to the intermediate frequency, this section of
the receiver may be designed for optimum selectivity and high amplification. High amplification is easily obtained in the intermediatefrequency amplifier, since it operates at a
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HANDBOOK
The
advantage over the tuned
radio frequency (t -r -f) type of receiver because
of what is commonly known as arithmetical
selectivity.
This can best be illustrated by considering
two receivers, one of the t -r -f type and one of
the superheterodyne type, both attempting to
PENTODE
1
To
BY -PASS
MEG.
CAPACITORS 05 TO 0.1 JIPD.
A V C
Figure
6
TYPICAL I -F AMPLIFIER STAGE
relatively low frequency, where conventional
pentode-type tubes give adequate voltage gain.
A typical i -f amplifier is shown in figure 6.
From the diagram it may be seen that both
the grid and plate circuits are tuned. The tuned
circuits used for coupling between i -f stages
are known as i-I transformers. These will be
more fully discussed later in this chapter.
Choice of Intermediate Frequency
The choice of a frequency
for the i -f amplifier involves several considera-
tions. One of these considerations concerns
selectivity; the lower the intermediate frequency the greater the obtainable selectivity.
On the other hand, a rather high intermediate
frequency is desirable from the standpoint of
image elimination, and also for the reception
of signals from television and FM transmitters
and modulated self -controlled oscillators, all
of which occupy a rather wide band of frequencies, making a broad selectivity characteristic
desirable. Images are a pecularity common to
all superheterodyne receivers, and for this
reason they are given a detailed discussion
later in this chapter.
While intermediate frequencies as low as
50 kc. are used where extreme selectivity is
a requirement, and frequencies of 60 Mc. and
above are used in some specialized forms of
receivers, most present -day communications
superheterodynes use intermediate frequencies
around either 455 kc. or 1600 kc.
Home -type broadcast receivers almost always use an i -f in the vicinity of 455 kc.,
while auto receivers usually use a frequency
of about 262 kc. The standard frequency for
the i -f channel of FM receivers is 10.7 Mc.
Television receivers use an i -f which covers
the band between about 21.5 and 27 Mc., although a new band between 41 and 46 Mc. is
coming into more common usage.
Arithmetical
Selectivity
209
an overwhelming
VARIABLE-1/
INPIp
Superhetrodyne
Aside from allowing the use of
fixed -tuned band -pass amplifier
stages, the superheterodyne has
receive a desired signal at 10,000 kc. and
eliminate a strong interfering signal at 10,010
kc. In the t -r -f receiver, separating these two
signals in the tuning circuits is practically
impossible, since they differ in frequency by
only 0.1 per cent. However, in a superheterodyne with an intermediate frequency of, for example, 1000 kc., the desired signal will be
converted to a frequency of 1000 kc. and the
interfering signal will be converted to a frequency of 1010 kc., both signals appearing at
the input of the i -f amplifier. In this case, the
two signals may be separated much more readily, since they differ by 1 per cent, or 10 times
as much as in the first case.
The converter stage, or mixer,
of a superheterodyne receiver
can be either one of two types:
(1) it may use a single envelope converter
tube, such as a 6K8, 6SA7, or 6BE6, or (2) it
may use two tubes, or two sets of elements in
the same envelope, in an oscillator -mixer arrangement. Figure 7 shows a group of circuits
of both types to illustrate present practice
with regard to types of converter stages.
Converter tube combinations such as shown
in figures 7A and 7B are relatively simple and
inexpensive, and they do an adequate job for
most applications. With a converter tube such
as the 6SB7 -Y or the 6BA7 quite satisfactory
performance may be obtained for the reception
of relatively strong signals (as for example
FM broadcast reception) up to frequencies in
excess of 100 Mc. However, the equivalent input noise resistance of such tubes is of the
order of 200,000 ohms, which is a rather high
value indeed. So such tubes are not suited for
operation without an r -f stage in the high frequency range if weak -signal reception is
The Converter
Stage
desired.
The 6L7 mixer circuit shown in figure 7C,
and the 6BA7 circuit of figure 71), also are
characterized by an equivalent input noise re-
sistance of several hundred thousand ohms, so
that these also must be preceded by one or
more r-f stages with a fairly high gain per
stage if a low noise factor is desired of the
complete receiver.
However, the circuit arrangements shown
at figures 7F and 6F are capable of low -noise
operation within themselves, so that these
circuits may be fed directly from the antenna
without an r -f stage and still provide a good
noise factor to the complete receiver. Note
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210
Radio Receiver
THE
Fundamentals
RADIO
6SÁ7, 6SB7Y,
roar
AMP
6BE6. 6BÁ7
+250
2
V
ULF
+ 50 V.
Figure
7
TYPICAL FREQUENCY- CONVERTER (MIXER) STAGES
The relative advantages of the different circuits are discussed in the text
that both these circuits use control -grid injection of both the incoming signal and the
local- oscillator voltage. Hence, paradoxically,
circuits such as these should be preceded by
an r -f stage if local- oscillator radiation is to
be held to any reasonable value of field intensity.
As the frequency of operation of
a superheterodyne receiver is increased above a few hundred megacycles the
signal -to -noise ratio appearing in the plate
circuit of the mixer tube when triodes or pentodes are employed drops to a prohibitively
low value. At frequencies above the upper -fre-
quency limit for conventional mixer stages,
mixers of the diode type are most commonly
employed. The diode may be either a vacuum tube heater diode of a special u -h -f design
such as the 9005, or it may be a crystal diode
of the general type of the 1N21 through 1N28
series.
Diode Mixers
12 -4
Mixer Noise'
and Images
The effects of mixer noise and images are
troubles common to all superheterodynes. Since
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HANDBOOK
Mixer Characteristics
both these effects can largely be obviated by
the same remedy, they will be considered together.
Mixer Noise
Mixer noise of the shot- effect
type, which is evidenced by a
hiss in the audio output of the receiver, is
caused by small irregularities in the plate current in the mixer stage and will mask weak
signals. Noise of an identical nature is generated in an amplifier stage, but due to the
fact that the conductance in the mixer stage
is considerably lower than in an amplifier
stage using the same tube, the proportion of
inherent noise present in a mixer usually is
considerably greater than in an amplifier stage
using a comparable tube.
Although this noise cannot be eliminated,
its effects can be greatly minimized by placing sufficient signal- frequency amplification
having a high signal -to -noise ratio ahead of
the mixer. This remedy causes the signal output from the mixer to be large in proportion to
the noise generated in the mixer stage. Increasing the gain after the mixer will be of no
advantage in eliminating mixer noise difficulties; greater selectivity after the mixer will
help to a certain extent, but cannot be carried
too far, since this type of selectivity decreases
the i -f band -pass and if carried too far will
not pass the sidebands that are an essential
part of a voice -modulated signal.
A triode having a high trans conductance is the quietest
mixer tube, exhibiting somewhat less gain but
a better signal -to -noise ratio than a comparable multi -grid mixer tube. However, below 30
Mc. it is possible to construct a receiver that
will get down to the atmospheric noise level
without resorting to a triode mixer. The additional difficulties experienced in avoiding
pulling, undesirable feedback, etc., when using
a triode with control -grid injection tend to make
multi -grid tubes the popular choice for this
application on the lower frequencies.
On very high frequencies, where set noise
rather than atmospheric noise limits the weak
signal response, triode mixers are more widely
used. A 6J6 miniature twin triode with grids
in push -pull and plates in parallel makes an
excellent mixer up to about 600 Mc.
Triode Mixers
The amplitude of the injection volt age will affect the conversion trans conductance of the mixer, and therefore should be made optimum if maximum signal -to -noise ratio is desired. If fixed bias is
employed on the injection grid, the optimum
injection voltage is quite critical. If cathode
bias is used, the optimum voltage is not so
critical; and if grid leak bias is employed, the
Injection
Voltage
211
optimum injection voltage is not at all critical
just so it is adequate. Typical optimum injection voltages will run from 1 to 10 volts for
control grid injection, and 45 volts or so for
screen or suppressor grid injection.
There always are two signal frequencies which will combine with a given
frequency to produce the same difference frequency. For example: assume a superheterodyne with its oscillator operating on a higher
frequency than the signal, which is common
practice in present superheterodynes, tuned to
receive a signal at 14,100 kc. Assuming an
i -f amplifier frequency of 450 kc., the mixer
input circuit will be tuned to 14,100 kc., and
the oscillator to 14,100 plus 450, or 14,550 kc.
Now, a strong signal at the oscillator frequency plus the intermediate frequency (14,550
plus 450, or 15,000 kc.) will also give a difference frequency of 450 kc. in the mixer out put and will be heard also. Note that the image
is always twice the intermediate frequency
away from the desired signal. Images cause
repeat points on the tuning dial.
The only way that the image could be eliminated in this particular case would be to make
the selectivity of the mixer input circuit, and
any circuits preceding it, great enough so that
the 15,000-kc. signal never reaches the mixer
grid in sufficient amplitude to produce interference.
For any particular intermediate frequency,
image interference troubles become increasingly greater as the frequency to which the
signal- frequency portion of the receiver is
tuned is increased. This is due to the fact that
the percentage difference between the desired
frequency and the image frequency decreases
as the receiver is tuned to a higher frequency.
The ratio of strength between a signal at the
image frequency and a signal at the frequency
to which the receiver is tuned producing equal
output is known as the image ratio. The higher
this ratio, the better the receiver in regard to
image- interference troubles.
kith but a single tuned circuit between the
mixer grid and the antenna, and with 400 -500
kc. i -f amplifiers, image ratios of 60 db and
over are easily obtainable up to frequencies
around 2000 kc. Above this frequency, greater
selectivity in the mixer grid circuit through
the use of additional tuned circuits between
the mixer and the antenna is necessary if a
good image ratio is to be maintained.
Images
12 -5
Z
-F Stages
Since the necessLry tuned circuits between
the mixer and the antenna can be combined
with tubes to form r -f amplifier stages, the
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Radio
212
Receiver Fundamentals
PENTODE
INPUT
6AB4,
6J6,
6J4,
12AT7
O
GROUNDED-GRID
C
RADIO
THE
-i
70
Figure
TYPICAL PENTODE
R -F
+120v.
8
AMPLIFIER STAGE
CATHODE- COUPLED
reduction of the effects of mixer noise and the
increasing of the image ratio can be accomplished in a single section of the receiver.
When incorporated in the receiver, this section is known simply as an r -/ amplifier; when
it is a separate unit with a separate tuning
control it is often known as a preselector.
Either one or two stages are commonly used
in the preselector or r -f amplifier. Some pre selectors use regeneration to obtain still
greater amplification and selectivity. An r -f
amplifier or preselector embodying more than
two stages rarely ever is employed since two
stages will ordinarily give adequate gain to
override mixer noise.
Generally speaking, atmospheric noise in the frequency
range above 30 Mc. is quite
low -so low, in fact, that the noise generated
within the receiver itself is greater than the
noise received on the antenna. Hence it is of
the greatest importance that internally generated noise be held to a minimum in a receiver.
At frequencies much above 300 Mc. there is
not too much that can be done at the present
state of the art in the direction of reducing
receiver noise below that generated in the converter stage. But in the v -h -f range, between
30 and 300 Mc., the receiver noise factor in a
well designed unit is determined by the characteristics of the first r -f stage.
The usual v -h -f receiver, whether for communications or for FM or TV reception, uses
a miniature pentode for the first r -f amplifier
stage. The 6AK5 is the best of presently available types, with the 6CB6 and the 6DC6 closely approaching the 6AK5 in performance. But
when gain in the first r -f stage is not so important, and the best noise factor must be obtained, the first r -f stage usually uses a triode.
Shown in figure 9 are four commonly used
types of triode r -f stages for use in the v -h -f
range. The circuit at (A) uses few components
and gives a moderate amount of gain with very
low noise. It is most satisfactory when the
first r -f stage is to be fed directly from a low-
6J6
+120
V
LOW NOISE
CASCODE
Lo
R -F
Stages in
the V -H -F Range
+120
V.
6BK7,6B07AOR6BZ7
200V.
Figure 9
TYPICAL TRIODE V -H -F
R -F AMPLIFIER STAGES
Triode r -f stages contribute the least amount of
noise output for a given signal level, hence their
frequent use in the v-h -f range.
impedance coaxial transmission line. Figure
9 (B) gives somewhat more gain than (A), but
requires an input matching circuit. The effective gain of this circuit is somewhat reduced
when it is being used to amplify a broad band
of frequencies since the effective Gm of the
cathode -coupled dual tube is somewhat less
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HANDBOOK
The
IA MC.
Cascode
R F.
AMPLIFIER
10 MC.
TUNABLE
MIXER
213
455 KC.
MC.
TUNABLE
Amplifier
DULATOR
FII%
MIXER
I.F.
AMPLIFIER
AMPLIFIER
CRYSTAL
OSCILLATOR
VARIABLE
OSCILLATOR
AND
AUDIO
3545 KC.
1
455 KC.
MIXER
50 KC.
I I
nx
I I
MIXER
I.I.
AMPLIFIER
l
I
L
FIXED
DEMODULATOR
AMPLIFIER
AND
AUDIO
II
I I
1A4S5KC
VARIABLE
FIXED
11
OSCILLATOR
OSCILLATOR
I
CONVENTIONAL COMMUNICATIONS
RECEIVER
505 BC.
I
II HIGHLY SELECTIVE ACCESSORY
II AMPLIFIER AND DEMODULATOR
I
F.
(Q5'ER)I
_JL
Figure
10
TYPICAL DOUBLE -CONVERSION SUPERHETERODYNE RECEIVERS
Illustrated at (A) is the basic circuit of a commercial double- conversion superheterodyne receiver. At (B) is
illustrated the application of on accessory sharp i -f channel for obtaining improved selectivity from a conventional communications receiver through the use of the double-conversion principle.
than half the
taken alone.
Gm
of either of the two tubes
The Cascode r -f amplifier, developed at the MIT Radiation
Laboratory during World War II,
is a low noise circuit employing a grounded
cathode triode driving a grounded grid triode,
as shown in figure 9C. The stage gain of such
a circuit is about equal to that of a pentode
tube, while the noise figure remains at the low
level of a triode tube. Neutralization of the
first triode tube is usually unnecessary below
50 Mc. Above this frequency, a definite improvement in the noise figure may be obtained
Through the use of neutralization. The neutralizing coil, LN, should resonate at the operating frequency with the grid -plate capacity of
the first triode tube.
The 6B(27A and 6BZ7 tubes are designed for
use in cascode circuits, and may be used to
good advantage in the 144 Mc. and 220 Mc. amateur bands (figure 9D). For operation at higher
frequencies, the 6A)4 tube is recommended.
The Cascode
Amplifier
As previously mentioned,
the use of a higher intermediate frequency will also improve the image
Double Conversion
ratio, at the expense of i -f selectivity, by
placing the desired signal and the image farther apart. To give both good image ratio at
the higher frequencies and good selectivity in
the i -f amplifier, a system known as double
conversion is sometimes employed. In this system, the incoming signal is first converted to
a rather high intermediate frequency, and then
amplified and again converted, this time to a
much lower frequency. The first intermediate
frequency supplies the necessary wide separation between the image and the desired signal, while the second one supplies the bulk of
the i -f selectivity.
The double -conversion system, as illustrated in figure 10, is receiving two general
types of application at the present time. The
first application is for the purpose of attaining
extremely good stability in a communications
receiver through the use of crystal control of
the first oscillator. In such an arrangement,
as used in several types of Collins receivers,
the first oscillator is crystal controlled and is
followed by a tunable i -f amplifier which then
is followed by a mixer stage and a fixed-tuned
i -f a m p l i f i e r on a touch lower frequency.
Through such a circuit arrangement the sta-
bility of the complete receiver is equal to the
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214
Radio
Receiver
THE
Fundamentals
RADIO
stability of the oscillator which feeds the second mixer, while the selectivity is determined
by the bandwidth of the second, fixed i -f am-
plifier.
The second common application of the
double- conversion principle is for the purpose
of obtaining a very high degree of selectivity
in the complete communications receiver. In
this type of application, as illustrated in figure 10 (B), a conventional communications receiver is modified in such a manner that its
normal i -f amplifier (which usually is in the
450 to 915 kc. range) instead of being fed to
a demodulator and then to the audio system,
is alternatively fed to a fixed -tune mixer stage
and then into a much lower intermediate frequency amplifier before the signal is demodulated and fed to the audio system. The accessory i -f amplifier system (sometimes called a
Q5'er) normally is operated on a frequency of
175 kc., 85 kc., or 50 kc.
12 -6
Signal- Frequency
Tuned Circuits
The signal- frequency tuned circuits in high frequency superheterodynes and tuned radio
frequency types of receivers consist of coils
of either the solenoid or universal -wound types
shunted by variable capacitors. It is in these
tuned circuits that the causes of success or
failure of a receiver often lie. The universal wound type coils usually are used at frequencies below 2000 kc.; above this frequency the
single-layer solenoid type of coil is more
satisfactory.
The two factors of greatest significance in determining the gain per -stage and selectivity, respectively, of a tuned amplifier are tuned- circuit
impedance and tuned -circuit Q. Since the resistance of modern capacitors is low at ordinary frequencies, the resistance usually can
be considered to be concentrated in the coil.
The resistance to be considered in making Q
determinations is the r -f resistance, not the
d -c resistance of the wire in the coil. The latter ordinarily is low enough that it may be
neglected. The increase in r -f resistance over
d -c resistance primarily is due to skin effect
and is influenced by such factors as wire size
and type, and the proximity of metallic objects
or poor insulators, such as coil forms with
high losses. Higher values of Q lead to better
selectivity and increased r -f voltage across
the tuned circuit. The increase in voltage is
due to an increase in the circuit impedance
with the higher values of Q.
Impedance
and Q
R F
C
INPUT
Figure 11
ILLUSTRATING "COMMON POINT"
BY- PASSING
To reduce the detrimental effects of cathode circuit inductance in v -h -f stages, all by -pass capacitors should be returned to the cathode terminal
at the socket. Tubes with two cathode leads can
give improved performance if the grid return is
made to one cathode terminal while the plate and
screen by -pass returns are made to the cathode
terminal which is connected to the suppressor
within the tube.
Frequently it is possible to secure an increase in impedance in a resonant circuit, and
consequently an increase in gain from an amplifier stage, by increasing the reactance
through the use of larger coils and smaller
tuning capacitors (higher L/C ratio).
Another factor which influences the operation of
tuned circuits is the input resistance of the
tubes placed across these circuits. At broadcast frequencies, the input resistance of most
conventional r -f amplifier tubes is high enough
so that it is not bothersome. But as the frequency is increased, the input resistance becomes lower and lower, until it ultimately
reaches a value so low that no amplification
can be obtained from the r -f stage.
The two contributing factors to the decrease
in input resistance with increasing frequency
are the transit time required by an electron
traveling between the cathode and grid, and
the inductance of the cathode lead common
to both the plate and grid circuits. As the
frequency becomes higher, the transit time
can become an appreciable portion of the time
required by an r -f cycle of the signal voltage,
and current will actually flow into the grid.
The result of this effect is similar to that
which would be obtained by placing a resistance between the tube's grid and cathode.
Input Resistance
Because the oscillator in a
superheterodyne operate s
"offset" from the other front
end circuits, it is necessary to make special
provisions to allow the oscillator to track
Superheterodyne
Tracking
when
similar tuning capacitor sections are
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Tuning Circuits
HANDBOOK
215
MIXER
PADDING CAPACITOR
TUNING CAPACITOR
OSCILLATOR
SERIES TRACKING CAPACITOR
Figure 13
BANDSPREAD CIRCUITS
Parallel bandspread is illustrated at (A) and (B),
series bandspread at (C), and tq,ped.coil band-
Figure 12
SERIES TRACKING EMPLOYED
IN THE H -F OSCILLATOR OF A
SUPERHETERODYNE
The series tracking capacitor permits the use of
identical gangs in a ganged capacitor, since the
tracking capacitor slows down the rate of frequency change in the oscillator so that a constant difference in frequency between the oscillator and
the r -f stage (equal to the i -f amplifier frequency)
may be maintained.
ganged. The usual method of obtaining good
tracking is to operate the oscillator on the
high- frequency side of the mixer and use a
series tracking capacitor to slow down the
tuning rate of the oscillator. The oscillator
tuning rate must be slower because it covers
a smaller range than does the mixer when both
are expressed as a percentage of frequency.
At frequencies above 7000 kc. and with ordinary intermediate frequencies, the difference
in percentage between the two tuning ranges
is so small that it may be disregarded in receivers designed to cover only a small range,
such as an amateur band.
A mixer and oscillator tuning arrangement
in which a series tracking capacitor is provided is shown in figure 12. The value of the
tracking capacitor varies considerably with
different intermediate frequencies and tuning
ranges, capacitances as low as .0001 pfd.
being used at the lower tuning -range frequencies, and values up to .01 µfd. being used at
the higher frequencies.
Superheterodyne receivers designed to cover
only a single frequency range, such as the
standard broadcast band, sometimes obtain
tracking between the oscillator and the r -f circuits by cutting the variable plates of the oscillator tuning section to a different shape
from those used to tune the r -f stages.
frequency to which a
receiver responds may be
varied by changing the size
of either the coils or the capacitors in the tuning circuits, or both. In short -wave receivers
Frequency Range
Selection
The
spread at (D),
combination of both methods is usually employed, the coils being changed from one band
to another, and variable capacitors being used
to tune the receiver across each band. In practical receivers, coils may be changed by one
of two methods: a switch, controllable from
the panel, may be used to switch coils of different sizes into the tuning circuits or, alternatively, coils of different sizes may be
plugged manually into the receiver, the connection into the tuning circuits being made by
suitable plugs on the coils. Where there are
several plug -in cods for each band, they are
sometimes arranged to a single mounting strip,
allowing them all to be plugged in simultaneously.
a
In receivers using large tuning
capacitors to cover the shortwave spectrum with a minimum
of coils, tuning is likely to be quite difficult,
owing to the large frequency range covered by
a small rotation of the variable capacitors.
To alleviate this condition, some method of
slowing down the tuning rate, or bandspread ing, must be used.
Bandspread
Tuning
Quantitatively, bandspread is usually designated as being inversely proportional to the
range covered. Thus, a large amount of bandspread indicates that a small frequency range
is covered by the bandspread control. Conversely, a small amount of bandspread is taken
to mean that a large frequency range is covered
by the bandspread dial.
Types of
Bandspread
Bandspreading systems are of
two general types: electrical and
mechanical. Mechanical systems
are exemplified by high -ratio dials in which
the tuning capacitors rotate much more slowly
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216
Radio
Receiver
THE
Fundamentals
than the dial knob. In this system, there is
often a separate scale or pointer either connected or geared to the dial knob to facilitate
accurate dial readings. However, there is a
practical limit to the amount of mechanical
bandspread which can be obtained in a dial
and capacitor before the speed- reduction unit
and capacitor bearings become prohibitively
expensive. Hence, most receivers employ a
combination of electrical and mechanical band spread. In such a system, a moderate reduction in the tuning rate is obtained in the dial,
and the rest of the reduction obtained by elec-
trical bandspreading.
In this book and in other radio
literature, mention is sometimes
made of stray or circuit capacitance. This capacitance is in the usual sense
defined as the capacitance remaining across
a coil when all the tuning, bandspread, and
padding capacitors across the circuit are at
their minimum capacitance setting.
Circuit capacitance can be attributed to two
general sources. One source is that due to the
input and output capacitance of the tube when
its cathode is heated. The input capacitance
varies somewhat from the static value when
the tube is in actual operation. Such factors
as plate load impedance, grid bias, and frequency will cause a change in input capacitance. However, in all except the extremely
high -transconductance tubes, the published
measured input capacitance is reasonably close
to the effective value when the tube is used
within its recommended frequency range. But
in the high -transconductance types the effective capacitance will vary considerably from
the published figures as operating conditions
are changed.
The second source of circuit capacitance,
and that which is more easily controllable, is
that contributed by the minimum capacitance
of the variable capacitors across the circuit
and that due to capacitance between the wiring and ground. In well -designed high -frequency receivers, every effort is made to keep
this portion of the circuit capacitance at a
minimum since a large capacitance reduces
the tuning range available with a given coil
and prevents a good L/C ratio, and consequently a high- impedance tuned circuit, from
being obtained.
A good percentage of stray circuit capacitance is due also to distributed capacitance
of the coil and capacitance between wiring
Stray Circuit
Capacitance
points and chassis.
Typical values of circuit capacitance may
run from 10 to 75 µpfd. in high- frequency re-
ceivers, the first figure representing concentric-line receivers with acorn or miniature
tubes and extremely small tuning capacitors,
RADIO
latter representing all -wave sets with
bandswitching, large tuning capacitors, and
conventional tubes.
and the
12 -7
I
-F Tuned Circuits
I -f amplifiers usually employ bandpass circuits of some sort. A bandpass circuit is exactly what the name implies -a circuit for passing a band of frequencies. Bandpass arrange-
ments can be designed for almost any degree
of selectivity, the type used in any particular
case depending upon the ultimate application
of the amplifier.
I.F
Transformers
Intermediate frequency trans formers ordinarily consist of
two or more tuned circuits and
some method of coupling the tuned circuits
together. Some representative arrangements
are shown in figure 14. The circuit shown at
A is the conventional i -f transformer, with the
coupling, M, between the tuned circuits being
provided by inductive coupling from one coil
to the other. As the coupling is increased, the
selectivity curve becomes less peaked, and
when a condition known as critical coupling
is reached, the top of the curve begins to flatten out. When the coupling is increased still
more, a dip occurs in the top of the curve.
The windings for this type of i -f transformer,
as well as most others, nearly always consist
of small, flat universal -wound pies mounted
either on a piece of dowel to provide an air
core or on powdered -iron for iron core i-f transformers. The iron -core transformers generally
have somewhat more gain and better selectivity
than equivalent air-core units.
The circuits shown at figure 14 -B and C are
quite similar. Their only difference is the type
of mutual coupling used, an inductance being
used at B and a capacitance at C. The operation of both circuits is similar. Three resonant circuits are formed by the components. In
B, for example, one resonant circuit is formed
by
C1, C, and L2 all in series. The frequency of this resonant circuit is just the same
as that of a single one of the coils and capacitors, since the coils and capacitors are similar in both sides of the circuit, and the resonant frequency of the two capacitors and the
two coils all in series is the same as that of
a single coil and capacitor. The second resonant frequency of the complete circuit is determined by the characteristics of each half of
the circuit containing the mutual coupling device. In B, this second frequency will be lower
than the first, since the resonant frequency of
C, and the inductance, M, or L2, C, and M
is lower than that of a single coil and capaci-
L
L
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HANDBOOK
I
tor, due to the inductance of M being added to
the circuit.
The opposite effect takes place at figure
14 -C, where the common coupling impedance
is a capacitor. Thus, at C the second resonant frequency is higher than the first. In either
case, however, the circuit has two resonant frequencies, resulting in a flat- topped selectivity
curve. The width of the top of the curve is
controlled by the reactance of the mutual
coupling component. As this reactance is increased (inductance made greater, capacitance
made smaller), the two resonant frequencies
become further apart and the curve is broadened.
In the circuit of figure 14 -D, there is inductive coupling between the center coil and each
of the outer coils. The result of this arrangement is that the center coil acts as a sharply
tuned coupler between the other two. A signal
somewhat off the resonant frequency of the
transformer will not induce as much current
in the center coil as will a signal of the correct frequency. When a smaller current is induced in the center coil, it in turn transfers
a still smaller current to the output coil. The
effective coupling between the outer coils increases as the resonant frequency is approached, and remains nearly constant over a
small range and then decreases again as the
resonant band is passed.
Another very satisfactory bandpass arrangement, which gives a very straight- sided, flat topped curve, is the negative- mutual arrangement shown at figure 14 -E. Energy is transferred between the input and output circuits in
this arrangement by both the negative- mutual
coils, M, and the common capacitive reactance,
C. The negative- mutual coils are interwound
on the same form, and connected backward.
Transformers usually are made tunable over
a small range to permit accurate alignment in
the circuit in which they are employed. This
is accomplished either by means of a variable
capacitor across a fixed inductance, or by
means of a fixed capacitor across a variable
inductance. The former usually employ either
mica -compression capacitor (designated
a
"mica tuned "), or a small air dielectric variable capacitor (designated "air tuned"). Those
which use a fixed capacitor usually employ a
powdered iron core on a threaded rod to vary
the inductance, and are known as "permea-
bility tuned."
It is obvious that to pass modulation sidebands and to allow
for slight drifting of the transmitter carrier frequency and the receiver local oscillator, the
i -f amplifier must pass not a single frequency
but a band of frequencies. The width of this
pass band, usually 5 to 8 kc. at maximum
Shape Factor
-F
M
Amplifiers
217
M
E
Figure 14
-F AMPLIFIER COUPLING
ARRANGEMENTS
The interstoge coupling arrangements illustrated
above give a better shape factor (more straight
sided selectivity curve) than would the some number of tuned circuits coupled by means of tubes.
I
width in
a good communications receiver, is
known as the pass band, and is arbitrarily
taken as the width between the two frequencies at which the response is attenuated 6 db,
or is "6 db down." However, it is apparent
that to discriminate against an interfering signal which is stronger than the desired signal,
much more than 6 db attenuation is required.
The attenuation arbitrarily taken to indicate
adequate discrimination against an interfering
signal is 60 db.
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218
Radio
Receiver
THE
Fundamentals
L
R
C
Figure
RADIO
16
ELECTRICAL EQUIVALENT OF
QUARTZ FILTER CRYSTAL
The crystal is equivalent to o very large value
of
inductance in series with small values of capacitance and resistance, with a larger though still
small value of capacitance across the whole circuit Yrepresenting holder capacitance plus stray
capacitances).
Figure 15
-F PASS BAND OF TYPICAL
COMMUNICATIONS RECEIVER
I
It is apparent that it is desirable to have
bandwidth at 60 db down as narrow as
possible, but it must be done without making
the pass band (6 db points) too narrow for satisfactory reception of the desired signal. The
figure of merit used to show the ratio of bandwidth at 6 db down to that at 60 db down is
designated shape factor. The ideal i -f curve,
a rectangle, would have a shape factor of 1.0.
The i -f shape factor in typical communications
receivers runs from 3.0 to 5.5.
The most practicable method of obtaining a
low shape factor for a given number of tuned
circuits is to employ them in pairs, as in figure 14 -A, adjusted to critical coupling (the
value at which two resonance points just begin to become apparent). If this gives too
sharp a "nose" or pass band, then coils of
lower Q should be employed, with the coupling
maintained at the critical value. As the Q is
lowered, closer coupling will be required for
critical coupling.
Conversely if the pass band is too broad,
coils of higher Q should be employed, the
coupling being maintained at critical. If the
pass band is made more narrow by using looser
coupling instead of raising the Q and main taninig critical coupling, the shape factor will
not be as good.
The pass band will not be much narrower
for several pairs of identical, critically coupled
tuned circuits than for a single pair. However,
the shape factor will be greatly improved as
each additional pair is added, up to about 5
pairs, beyond which the improvement for each
additional pair is not significant. Commercially available communications receivers of
the
good quality normally employ
3 or 4 double
tuned transformers with coupling adjusted to
critical or slightly less.
The pass band of a typical communication
receiver having a 455 kc. i -f amplifier is shown
in figure 15.
"Miller
Effect"
As mentioned previously, the dynamic input capacitance of a tube varies
slightly with bias. As a -v-c voltage
normally is applied to i -f tubes for radiotelephony reception, the effective grid-cathode
capacitance varies as the signal strength
varies, which produces the same effect as
slight detuning of the i -f transformer. This
effect is known as "Miller effect," and can
be minimized to the extent that it is not
troublesome either by using a fairly low L/C
ratio in the transformers or by incorporating
a small amount of degenerative feedback, the
latter being most easily accomplished by leaving part of the cathode resistor unbypassed
for r.f.
Crystal Filters
The pass band of an intermediate frequency amplifier
may be made very narrow through the use of a
piezoelectric filter crystal employed as a
series resonant circuit in a bridge arrangement known as a crystal filter. The shape factor is quite poor, as would be expected when
the selectivity is obtained from the equivalent
of a single tuned circuit, but the very narrow
pass band obtainable as a result of the extremely high Q of the crystal makes the crystal filter useful for c -w telegraphy reception.
The pass band of a 455 kc. crystal filter may
be made as narrow as 50 cycles, while the
narrowest pass band that can be obtained with
a 455 kc. tuned circuit of practicable dimensions is about 5 kc.
The electrical equivalent of a filter crystal
is shown in figure 16. For a given frequency,
L is very high, C very low, and R (assuming
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Filters
Crystal
HANDBOOK
219
CRYSTAL
E
SELECTIVITY
CONTROL
PHASING
CONTROL
Figure
For
a
Figure 17
EQUIVALENT OF CRYSTAL
FILTER CIRCUIT
given voltage out of the generator, the volt-
age developed across Z1 depends upon the ratio
of the impedance of X to the sum of the impedances
of Z and Z1. Because of the high
of the crystal,
its impedance changes rapidly with changes in
Q
frequency.
a good crystal of high Q) is very low. Capacitance C, represents the shunt capacitance of
the electrodes, plus the crystal holder and
wiring, and is many times the capacitance of
C. This makes the crystal act as a parallel
resonant circuit with a frequency only slightly
higher than that of its frequency of series
resonance. For crystal filter use it is the
series resonant characteristic that we are primarily interested in.
The electrical equivalent of the basic crystal filter circuit is shown in figure 17. If the
impedance of Z plus Z, is low compared to the
impedance of the crystal X at resonance, then
the current flowing through
and the voltage
developed across it, will be almost in inverse
proportion to the impedance of X, which has
a very sharp resonance curve.
If the impedance of Z plus Z, is made high
compared to the resonant impedance of X, then
there will be no appreciable drop in voltage
across Z, as the frequency departs from the
resonant frequency of X until the point is
reached where the impedance of X approaches
that of Z plus Z,. This has the effect of broadening out the curve of frequency versus voltage
developed across
which is another way of
saying that the selectivity of the crystal filter
(but not the crystal proper) has been reduced.
In practicable filter circuits the impedances
Z and Z, usually are represented by some form
of tuned circuit, but the basic principle of
operation is the same.
Z
Z
Fractical Filters
It is necessary to balance
out the capacitance across
the crystal holder (C in figure 16) to prevent
bypassing around the crystal undesired signals
off the crystal resonant frequency. The balancing is done by a phasing circuit which
takes out -of-phase voltage from a balanced in-
18
TYPICAL CRYSTAL FILTER CIRCUIT
put circuit and passes it to the output side of
the crystal in proper phase to neutralize that
passed through the holder capacitance. A rep-
resentative practical filter arrangement is
shown in figure 18. The balanced input circuit
may be obtained either through the use of a
split- stator capacitor as shown, or by the use
of a center -tapped input coil.
Variable- Selec-
circuit of figure 18, the
selectivity is minimum with
the crystal input circuit tuned
to resonance, since at resonance the impedance of the tuned circuit is maximum. As the
input circuit is detuned from resonance, however, the impedance decreases, and the selectivity becomes greater. In this circuit, the output from the crystal filter is tapped down on
the i -f stage grid winding to provide a low
value of series impedance in the output circuit. It will be recalled that for maximum selectivity, the total impedance in series with the
crystal (both input and output circuits) must
be low. If one is made low and the other is
made variable, then the selectivity may be
tivity Filters
In the
varied at will from sharp to broad.
The circuit shown in figure 19 also achieves
variable selectivity by adding a variable impedance in series with the crystal circuit. In
this case, the variable impedance is in series
with the crystal output circuit. The impedance
of the output circuit is varied by varying the
Q. As the Q is reduced (by adding resistance
in series with the coil), the impedance decreases and the selectivity becomes greater.
The input circuit impedance is made low by
using a non -resonant secondary on the input
transformer.
A variation of the circuit shown at figure
19 consists of placing the variable resistance
across the coil and capacitor, rather than in
series with them. The result of adding the resistor is a reduction of the output impedance,
and an increase in selectivity. The circuit behaves oppositely to that of figure 19, however;
as the resistance is lowered the selectivity
becomes greater. Still another variation of figure 19 is to use the tuning capacitor across
the output coil to vary the output impedance.
www.americanradiohistory.com
220
Radio
Receiver Fundamentals
RADIO
THE
CRYSTwL
SELECTIVITY
CONTROL
Figure
r
CRYSTAL NOTCH
+I
+2 +3
z
3 3S
19
VARIABLE SELECTIVITY
CRYSTAL FILTER
O
This circuit permits of a greater control of selectivity than does the circuit of figure 16, and does
not require a split- stator variable capacitor.
m
0 40
J
V
CI
-3
-I
-2
so
455
+
KC
Figure 20
-F PASS BAND OF TYPICAL
CRYSTAL FILTER
COMMUNICATIONS RECEIVER
As the output circuit is detuned from resonance, its impedance is lowered, and the
selectivity increases. Sometimes a set of
fixed capacitors and a multipoint switch are
used to give step -by -step variation of the output circuit tuning, and thus of the crystal
filter selectivity.
As previously discussed, a filter
crystal has both a resonant(series
resonant) and an anti -resonant
(parallel resonant) frequency, the impedance
of the crystal being quite low at the former
frequency, and quite high at the latter frequency. The anti- resonant frequency is just
slightly higher than the resonant frequency,
the difference depending upon the effective
shunt capacitance of the filter crystal and
holder. As adjustment of the phasing capacitor
controls the effective shunt capacitance of the
crystal, it is possible to vary the anti -resonant frequency of the crystal slightly without
unbalancing the circuit sufficiently to let undesired signals leak through the shunt capacitance in appreciable amplitude. At the exact
anti -resonant frequency of the crystal the attenuation is exceedingly high, because of the
high impedance of the crystal at this frequency. This is called the rejection notch, and
can be utilized virtually to eliminate the
heterodyne image or repeat tuning of c -w signals. The beat frequency oscillator can be
so adjusted and the phasing capacitor so adjusted that the desired beat note is of such
a pitch that the image (the same audio note
on the other side of zero beat) falls in the rejection notch and is inaudible. The receiver
then is said to be adjusted for single -signal
Rejection
Notch
operation.
The rejection notch sometimes can be employed to reduce interference from an undesired phone signal which is very close in
frequency to a desired phone signal. The filter
is adjusted to "broad" so as to permit tele-
I
phony reception, and the receiver tuned so
that the carrier frequency of the undesired
signal falls in the rejection notch. The modulation sidebands of the undesired signal still
will come through, but the carrier heterodyne
will
be effectively eliminated and interference
greatly reduced.
A typical crystal selectivity curve for a
communications receiver is shown in figure 20.
Crystal Filter
Considerations
A
crystal
filter, especially
when adjusted for single sig-
nal reception, greatly reduces
interference and background noise, the latter
feature permitting signals to be copied that
would ordinarily be too weak to be heard above
the background hiss. However, when the filter
is adjusted for maximum selectivity, the pass
band is so narrow that the received signal
must have a high order of stability in order to
stay within the pass band. Likewise, the local
oscillator in the receiver must be highly stable,
or constant retuning will be required. Another
effect that will be noticed with the filter adjusted too "sharp" is a tendency for code
characters to produce a ringing sound, and
have a hangover or "tails." This effect limits
the code speed that can be copied satisfactorily when the filter is adjusted for extreme
selectivity.
The Collins Mechanical Fil ter (figure 21) is a new concept in the field of selectivity. It is an electro- mechanical bandpass
filter about half the size of a cigarette package. As shown in figure 22, it consists of an
input transducer, a resonant mechanical secThe Mechanical
Filter
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Collins Mechanical Filter
HANDBOOK
tion comprised of a number of metal discs, and
an output transducer.
The frequency characteristics of the resonant mechanical section provide the almost
rectangular selectivity curves shown in figure
23. The input and output transducers serve
only as electrical to mechanical coupling devices and do not affect the selectivity characteristics which are determined by the metal
discs. An electrical signal applied to the input terminals is converted into a mechanical
vibration at the input transducer by means of
magnetostriction. This mechanical vibration
travels through the resonant mechanical section to the output transducer, where it is converted by magnetostriction to an electrical
signal which appears at the output terminals.
In order to provide the most efficient electromechanical coupling, a small magnet in the
mounting above each transducer applies a magnetic bias to the nickel transducer core. The
electrical impulses then add to or subtract
from this magnetic bias, causing vibration of
the filter elements that corresponds to the
exciting signal. There is no mechanical motion
except for the imperceptible vibration of the
metal discs.
Magnetostrictively -driven mechanical filters
have several advantages over electrical equivalents. In the region from 100 kc. to 500 kc.,
the mechanical elements are extremely small,
and a mechanical filter having better selectivity than the best of conventional i -f systems
may be enclosed in a package smaller than one
i -f transformer.
Since mechanical elements with Q's of 5000
or more are readily obtainable, mechanical filters may be designed in accordance with the
theory for lossless elements. This permits filter characteristics that are unobtainable with
electrical circuits because of the relatively
high losses in electrical elements as compared
with the mechanical elements used in the
221
c.
Figure 21
COLLINS MECHANICAL FILTERS
The
Collins Mechanical Filter is an
electro- mechanical bandpass filter which
surpasses, in one small unit, the selectivity of conventional, space-consuming
filters. At the left is the miniaturized
filter, less than 2!4' long. Type H is
next, and two horizontal mounting types
are at right. For exploded view of Collins
Mechanical Filter, see figure 46.
The frequency characteristics of the mechanical filter are permanent, and no adjustment is
required or is possible. The filter is enclosed
in a hermetically sealed case.
In order to realize full benefit from the mechanical filter's selectivity characteristics,
it is necessary to provide shielding between
the external input and output circuits, capable
of reducing transfer of energy external to the
o
filters.
ONE SUPPORTING
DISC AT
EACH END
RESONANT
MECHANICAL SECTION
(0 RESONANT DISCS)
ill
COUPLING RODS
DIAS MAGNET
iì%U
MAGNETOSTRICTIVE
DRIVING ROD
RANSDUCER
COIL
ELECTRICAL SIGNAL
(INPUT OR OUTPUT)
ELECTRICAL SIGNAL
(INPUT OR OUTPUT)
Figure 22
MECHANICAL FILTER
FUNCTIONAL DIAGRAM
Figure 23
Selectivity curves of 4554c. mechanical filters
with nominal 0.8 -%c. (dotted line) and 3.1 -kc.
(solid line) bandwidth at -6 db.
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222
Radio Receiver
THE
Fundamentals
I VERY SMALL
65J7
I.F. STAGE
RADIO
DET.
AUDIO
//1
Por
I-rlp 2Np
I
DETECTOR
BY
J
pA
I.F STAGE
Figure 24
A
GRID LEAK DETECTOR
DET.
VARIABLE -OUTPUT B -F -O CIRCUIT
beat- frequency oscillator whose output is con-
trollable is of considerable assistance in copying
c-w signals over a wide ronge of levels, and such
a control is almost a necessity for satisfactory
copying of single -sideband radiophone signals.
AUDIO
filter by a minimum value of 100 db. If the input circuit is allowed to couple energy into
the output circuit external to the filter, the
excellent skirt selectivity will deteriorate and
the passband characteristics will be distorted.
As with almost any mechanically resonant
circuit, elements of the mechanical filter have
multiple resonances. These result in spurious
modes of transmission through the filter and
produce minor passbands at frequencies on
other sides of the primary passband. Design
of the filter reduces these sub -bands to a low
level and removes them from the immediate
area of the major passband. Two conventional
i -f transformers supply increased attenuation
to these spurious responses, and are sufficient
to reduce them to an insignificant level.
The beat -frequency oscillator,
usually called the b.J.o., is a
necessary adjunct for reception of c -w telegraph signals on superheterodynes which have no other provision for obtaining modulation of an incoming c -w telegraphy signal. The oscillator is coupled into
or just ahead of second detector circuit and
supplies a signal of nearly the same frequency
as that of the desired signal from the i -f amplifier. If the i -f amplifier is tuned to 455 kc.,
for example, the b.f.o. is tuned to approximately 454 or 456 kc. to produce an audible
(1000 cycle) beat note in the output of the
second detector of the receiver. The carrier
signal itself is, of course, inaudible. The b.f.o.
is not used for voice reception, except as an
aid in searching for weak stations.
The b -f -o input to the second detector need
only be sufficient to give a good beat note on
an average signal. Too much coupling into the
second detector will give an excessively high
hiss level, masking weak signals by the high
noise background.
Figure 24 shows a method of manually ad-
OB
DIODE DETECTOR
I.F. STAGE
DET.
AUDIO
©
PLATE DETECTOR
I.F. STAGE
DET.
Beat- Frequency
Oscillators
OD
INFINITE IMPEDANCE DETECTOR
Figure 25
TYPICAL CIRCUITS FOR GRID -LEAK,
DIODE, PLATE AND INFINITE IMPEDANCE DETECTOR STAGES
justing the b -f -o output to correspond with the
strength of received signals. This type of variable b -f -o output control is a useful adjunct
to any superheterodyne, since it allows sufficient b-f-o output to be obtained to beat with
strong signals or to allow single - sideband
reception and at the same time permits the
b -f -o output, and consequently the hiss, to be
reduced when attempting to receive weak signals. The circuit shown is somewhat better
than those in which one of the electrode volt-
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HANDBOOK
Detector Circuits
TYPICAL
A -V -C
223
Figure 26
CIRCUIT USING
A DOUBLE DIODE
Any of the small dual-diode tubes may be used in
this circuit. Or, if desired, a duo- diodetriode may be used, with the triode acting as the first audio
stage. The left-hand diode serves
as the detector, while the right -hand side
acts as the a-v -c rectifier. The use of separate
diodes for detector and o-v -c reduces distortion when
receiving an AM signal with a high
modulation percentage.
b -f-o tube is changed, as the latter
circuits usually change the frequency of the
ages on the
b.f.o. at the same time they change the strength,
making it necessary to reset the trimmer each
time the output is adjusted.
The b.f.o. usually is provided with a small
trimmer which is adjustable from the front
panel to permit adjustment over a range of 5
or 10 kc. For single -signal reception the b.f.o.
always is adjusted to the high- frequency side,
in order to permit placing the heterodyne image
in the rejection notch.
In order to reduce the b -f -o signal output
voltage to a reasonable level which will prevent blocking the second detector, the signal
voltage is delivered through a low- capacitance
(high -reactance) capacitor having a value of
1 to 2 fgifd.
Care must be taken with the b.f.o. to prevent harmonics of the oscillator from being
picked up at multiples of the b -f-o frequency.
The complete b.f.o. together with the coupling
circuits to the second detector, should be thoroughly shielded to prevent pickup of the harmonics by the input end of the receiver.
If b-f-o harmonics still have a tendency to
give trouble after complete shielding and isolation of the b -f-o circuit has been accomplished, the passage of these harmonics from
the b-f-o circuit to the rest of the receiver can
be stopped through the use of a low -pass filter
in the lead between the output of the b -f -o circuit and the point on the receiver where the
b-f-o signal is to be injected.
12 -8
Detector, Audio, and
Control Circuits
Detectors
Second detectors for use in superheterodynes are usually of the
diode, plate, or infinite-impedance types. Occasionally, grid-leak detectors are used in receivers using one i-f stage or none at all, in
which case the second detector usually is
made regenerative.
Diodes are the most popular second detectors because they allow a simple method of
obtaining automatic volume control to be used.
Diodes load the tuned circuit to which they
are connected, however, and thus reduce the
selectivity slightly. Special i -f transformers
are used for the purpose of providing a low impedance input circuit to the diode detector.
Typical circuits for grid -leak, diode, plate
and infinite -impedance detectors are shown
in figure 25.
The elements of an automatic
volume control (a.v.c.) system are shown in figure 26.
A dual -diode tube is used as a combination
diode detector and a -v -c rectifier. The left hand diode operates as a simple rectifier in
the manner described earlier in this chapter.
Audio voltage, superimposed on a d -c voltage,
appears across the 500,000 -ohm potentiometer
(the volume control) and the .0001 -µfd. capacitor, and is passed on to the audio amplifier.
The right -hand diode receives signal voltage
directly from the primary of the last i -f amplifier, and acts as the a -v -c rectifier. The pulsating d-c voltage across the 1- megohm a.v.c.diode load resistor is filtered by a 500,000 -ohm
resistor and a .05-pfd. capacitor, and applied
as bias to the grids of the r -f and i -f amplifier
tubes; an increase or decrease in signal
strength will cause a corresponding increase
or decrease in a-v -c bias voltage, and thus the
gain of the receiver is automatically adjusted
to compensate for changes in signal strength.
Automatic Volurne Control
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224
Radio
A -C Loading of
Second Detector
Receiver
THE
Fundamentals
By disassociating the a.v.c.
and detecting functions
through using separate
diodes, as shown, most of the ill effects of
a-c shunt loading on the detector diode are
avoided. This type of loading causes serious
distortion, and the additional components required to eliminate it are well worth their cost.
Even with the circuit shown, a -c loading can
occur unless a very high (5 megohms, or more)
value of grid resistor is used in the following
audio amplifier stage.
A.V.C. in
IF
IF
RF
RADIO
RFoR IF
In receivers having
a beat frequency oscillator for the
reception of radiotelegraph
signals, the use of a.v.c.
can result in a great loss in sensitivity when
the b.f.o. is switched on. This is because the
beat oscillator output acts exactly like a
strong received signal, and causes the a -v -c
circuit to put high bias on the r -f and i -f stages,
thus greatly reducing the receiver's sensitivity. Due to the above effect, it is necessary to
provide a method of making the a -v -c circuit
inoperative when the b.f.o. is being used. The
simplest method of eliminating the a -v -c action is to short the a -v-c line to ground when
the b.f.o. is turned on. A two -circuit switch
may be used for the dual purpose of turning on
the beat oscillator and shorting out the a.v.c.
if desired.
B-F-O- Equipped
Receivers
Visual means for determining
whether or not the receiver is
properly tuned, as well as an
indication of the relative signal strength, are
both provided by means of tuning indicators
(S meters) of the meter or vacuum -tube type.
A d -c milliammeter can be connected in the
plate supply circuit of one or more r -f or i -f
amplifiers, as shown in figure 27A, so that the
change in plate current, due to the action of
the a -v -c voltage, will be indicated on the instrument. The d -c instrument MA should have
a full -scale reading approximately equal to the
total plate current taken by the stage or stages
whose plate current passes through the instrument. The value of this current can be estimated by assuming a plate current on each
stage (with no signal input to the receiver) of
Signal Strength
R
Fon
I
F
+70
6U5/6G5
OR
6E5
TO A V C
Indicators
about 6 ma. However, it will be found to be
more satisfactory to measure the actual plate
current on the stages with a milliammeter of
perhaps 0 -100 ma. full scale before purchasing
an instrument for use as an S meter. The
50 -ohm potentiometer shown in the drawing is
used to adjust the meter reading to full scale
with no signal input to the receiver.
When an ordinary meter is used in the plate
circuit of a stage, for the purpose of indicating signal strength, the meter reads backwards
t250v
Figure 27
SIGNAL -STRENGTH -METER CIRCUITS
Shown above are four circuits for obtaining a signal- strength reading which is a function of incoming carrier amplitude. The circuits are discussed
lin the accompanying text.
with respect to strength. This is because increased a -v -c bias on stronger signals causes
lower plate current through the meter. For this
reason, special meters which indicate zero at
the right -hand end of the scale are often used
for signal strength indicators in commercial
receivers using this type of circuit. Alternatively, the meter may be mounted upside
down, so that the needle moves toward the
right with increased strength.
The circuit of figure 27B can frequently be
used to advantage in a receiver where the cathode of one of the r-f or i -f amplifier stages
runs directly to ground through the cathode
bias resistor instead of running through a cath-
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HANDBOOK
ode -voltage gain control. In this case a 0 -1
d -c milliammeter in conjunction with a resistor
from 1000 to 3000 ohms can be used as shown
as a signal- strength meter. With this circuit
the meter will read backwards with increasing
signal strength as in the circuit previously
discussed.
Figure 27C is the circuit of a forward-readingS meter as is often used in communications
receivers. The instrument is used in an unbalanced bridge circuit with the d -c plate resistance of one i -f tube as one leg of the
bridge and with resistors for the other three
legs. The value of the resistor R must be determined by trial and error and will be somewhere in the vicinity of 50,000 ohms. Sometimes the screen circuits of the r -f and i -f
stages are taken from this point along with
the screen-circuit voltage divider.
Electron -ray tubes (sometimes called "magic
eyes") can also be used as indicators of relative signal strength in a circuit similar to that
shown in figure 27D. A 6U5/6G5 tube should
be used where the a-v -c voltage will be from
5 to 20 volts and a type 6E5 tube should be
used when the a-v-c voltage will run from 2
to 8 volts.
amplifiers are employed in nearly all radio
receivers. The audio amplifier stage or stages
are usually of the Class A type, although Class
AB push -pull stages are used in some receivers. The purpose of the audio amplifier is
to bring the relatively weak signal from the
detector up to a strength sufficient to operate
a pair of headphones or a loud speaker. Either
triodes, pentodes, or beam tetrodes may be
used, the pentodes and beam tetrodes usually
giving greater output. In some receivers, particularly those employing grid leak detection,
it is possible to operate the headphones directly from the detector, without audio amplification. In such receivers, a single audio
stage with a beam tetrode or pentode tube is
ordinarily used to drive the loud speaker.
Most communications receivers, either home constructed or factory -made, have a single ended beam tetrode (such as a 6L6 or 6V6) or
pentode (6F6 or 6K6 -GT) in the audio output
stage feeding the loudspeaker. If precautions
are not taken such a stage will actually bring
about a decrease in the effective signal -tonoise ratio of the receiver due to the rising
high- frequency characteristic of such a stage
when feeding a loud -speaker. One way of improving this condition is to place a mica or
paper capacitor of approximately 0.003 pfd.
capacitance across the primary of the output
transformer. The use of a capacitor in this
manner tends to make the load impedance seen
by the plate of the output tube more constant
Audio Ampl if iers
Audio
Noise
Suppression
225
over the audio -frequency range. The speaker
and transformer will tend to present a rising
impedance to the tube as the frequency increases, and the parallel capacitor will tend
to make the total impedance more constant
since it will tend to present a decreasing impedance with increasing audio frequency.
A still better way of improving the frequency
characteristic of the output stage, and at the
same time reducing the harmonic distortion,
is to use shunt feedback from the plate of the
output tube to the plate of a tube such as a
6SJ7 acting as an audio amplifier stage ahead
of the output stage.
Noise Suppression
12-9
The problem of noise suppression confronts
the listener who is located in places where
interference from power lines, electrical appliances, and automobile ignition systems is
troublesome. This noise is often of such intensity as to swamp out signals from desired
stations.
There are two principal methods for reducing this noise:
(1) A-c line filters at the source of interference, if the noise is created by an
electrical appliance.
(2) Noise -limiting circuits for the reduction, in the receiver itself, of interference of the type caused by automobile
ignition systems.
appliances, such
as electric mixers, heating pads,
vacuum sweepers, refrigerators,
oil burners, sewing machines, doorbells, etc.,
create an interference of an intermittent nature. The insertion of a line filter near the
source of interference often will effect a complete cure. Filters for small appliances can
consist of a 0.1 -µfd. capacitor connected across the 110 -volt a -c line. Two capacitors
in series across the line, with the midpoint
connected to ground, can be used in conjunction with ultraviolet ray machines, refrigerators, oil burner furnaces, and other more stubborn offenders. In severe cases of interference, additional filters in the form of heavy duty r -f choke coils must be connected in
series with the 110 -volt a -c line on both sides
of the line right at the interfering appliance.
Power Line
Filters
Many household
Numerous noise -limiting circuits
which are beneficial in overcoming key clicks, automobile ignition interference, and similar noise impulses
have become popular. They operate on the
Peak Noise
Limiters
principle that each individual noise pulse is
226
Radio
of very short duration, yet of very high amplitude. The popping or clicking type of noise
from electrical ignition systems may produce
a signal having a peak value ten to twenty
times as great as the incoming radio signal,
but an average power much less than the signal.
As the duration of this type of noise peak
is short, the receiver can be made inoperative
during the noise pulse without the human ear
detecting the total loss of signal. Some noise
limiters actually punch a bole in the signal,
while others merely limit the maximum peak
signal which reaches the headphones or loud-
speaker.
The noise peak is of such short duration
that it would not be objectionable except for
the fact that it produces an over -loading effect
on the receiver, which increases its time constant. A sharp voltage peak will give a kick
to the diaphragm of the headphones or speaker, and the momentum or inertia keeps the
diaphragm in motion until the dampening of
the diaphragm stops it. This movement produces a popping sound which may completely
obliterate the desired signal. If the noise pulse
can be limited to a peak amplitude equal to
that of the desired signal, the resulting interference is practically negligible for moderately low repetition rates, such as ignition
noise.
In addition, the i -f amplifier of the receiver
will also tend to lengthen the duration of the
noise pulses because the relatively high -Q i -f
tuned circuits will ring or oscillate when excited by a sharp pulse, such as produced by
ignition noise. The most effective noise limiter
would be placed before the high -Q i -f tuned
circuits. At this point the noise pulse is the
sharpest and has not been degraded by passage through the i -f transformers. In addition,
the pulse is eliminated before it can produce
ringing effects in the i -f chain.
noise limiter is shown in
figure 28. This is an adaptation of the Lamb noise silencer circuit. The i-f signal is fed into a double
grid tube, such as a 6L7, and thence into the
i -f chain. A 6AB7 high gain pentode is capacity coupled to the input of the i -f system.
This auxiliary tube amplifies both signal and
noise that is fed to it. It has a minimum of
selectivity ahead of it so that it receives the
true noise pulse before it is degraded by the
i -f strip. A broadly tuned i -f transformer is used
to couple the noise amplifier to a 6H6 noise
rectifier. The gain of the noise amplifier is
controlled by a potentiometer in the cathode
of the 6AB7 noise amplifier. This potentiometer controls the gain of the noise amplifier
The Lomb
Noise Limiter
THE
Receiver Fundamentals
An i -f
IST
DET
RADIO
ISTI.F.
2ND
I.F.
617
Figure 28
THE LAMB I -F NOISE SILENCER
stage and in addition sets the bias level on
the 6H6 diode so that the incoming signal will
not be rectified. Only noise peaks louder than
the signal can overcome the resting bias of
the 6H6 and cause it to conduct. A noise pulse
rectified by the 6H6 is applied as a negative
voltage to the control grid of the 6L7 i -f tube,
disabling the tube, and punching a hole in the
signal at the instant of the noise pulse. By
varying the bias control of the noise limiter,
the negative control voltage applied to the
6L7 may be adjusted until it is barely sufficient to overcome the noise impulses applied
to the al control grid without allowing the
modulation peaks of the carrier to become
badly distorted.
effective i -f noise
limiter is the Bishop limiter.
This is a full -wave shunt type
diode limiter applied to the primary of the last
i -f transformer of a receiver. The limiter is
self- biased and automatically adjusts itself
to the degree of modulation of the received
signal. The schematic of this limiter is shown
in figure 29. The bias circuit time constant is
determined by C, and the shunt resistance,
The Bishop
Another
Noise Limiter
which consists of R, and R2 in series. The
plate resistance of the last i -f tube and the
capacity of C, determine the charging rate of
the circuit. The limiter is disabled by opening
which allows the bias to rise to the value
of the i -f signal.
S
www.americanradiohistory.com
HANDBOOK
Noise Limiters
227
ter peak noise suppression than a standard
communications receiver having an i -f bandwidth of perhaps 8 kc. Likewise, when a crystal filter is used on the "sharp" position an
a -f peak' limiter is of little benefit.
Practical
Noise limiters range all the
way from an audio stage
Limiter Circuits
running at very low screen
or plate voltage, to elaborate affairs employing 5 or more tubes. Rather
than attempt to show the numerous types, many
of which are quite complex considering the
results obtained, only two very similar types
will be described. Either is just about as effective as the most elaborate limiter that can
be constructed, yet requires the addition of
but a single diode and a few resistors and
capacitors over what would be employed in a
good superheterodyne without a limiter. Both
circuits, with but minor modifications in resistance and capacitance values, are incorporated in one form or another in different
types of factory -built communications receivPeak Noise
Figure 29
THE BISHOP I -F NOISE LIMITER
Audio Noise
Limiters
of the simplest and most
practical peak limiters for radioSome
telephone reception employ one
or two diodes either as shunt or series limiters
in the audio system of the receiver. when a
noise pulse exceeds a certain predetermined
threshold value, the limiter diode acts either
as a short or open circuit, depending upon
whether it is used in a shunt or series circuit.
The threshold is made to occur at a level high
enough that it will not clip modulation peaks
enough to impair voice intelligibility, but low
enough to limit the noise peaks effectively.
Because the action of the peak limiter is
needed most on very weak signals, and these
usually are not strong enough to produce proper
a -v-c action, a threshold setting that is correct for a strong phone signal is not correct
for optimum limiting on very weak signals. For
this reason the threshold control often is tied
in with the a -v -c system so as to make the
optimum threshold adjustment automatic instead of manual.
Suppression of impulse noise by means of
an audio peak limiter is best accomplished at
the very front end of the audio system, and
for this reason the function of superheterodyne
second detector and limiter often are combined
in a composite circuit.
The amount of limiting that can be obtained
is a function of the audio distortion that can
be tolerated. Because excessive distortion
will reduce the intelligibility as much as will
background noise, the degree of limiting for
which the circuit is designed has to be a compromise.
Peak noise limiters working at the second
detector are much more effective when the i -f
bandwidth of the receiver is broad, because a
sharp i -f amplifier will lengthen the pulses by
the time they reach the second detector, making the limiter less effective. V-h -f super heterodynes have an i -f bandwidth considerably wider than the minimum necessary for
voice sidebands (to take care of drift and instability). Therefore, they are capable of bet-
ers.
Referring to figure 30, the first circuit shows
conventional superheterodyne second detector, a.v.c., and first audio stage with the
addition of one tube element,
which may
be either a separate diode or part of a twindiode as illustrated. Diode D, acts as a series
gate, allowing audio to get to the grid of the
a -f tube only so long as the diode is conducting. The diode is biased by a d -c voltage obtained in the same manner as a -v -c control voltage, the bias being such that pulses of short
duration no longer conduct when the pulse
voltage exceeds the carrier by approximately
60 per cent. This also clips voice modulation
peaks, but not enough to impair intelligibility.
It is apparent that the series diode clips
only positive modulation peaks, by limiting
upward modulation to about 60 per cent. Negative or downward peaks are limited automatically to 100 per cent in the detector, because
obviously the rectified voltage out of the
diode detector cannot be less than zero. Limiting the downward peaks to 60 per cent or so
instead of 100 per cent would result in but
little improvement in noise reduction, and the
results do not justify the additional components required.
It is important that the exact resistance
values shown be used, for best results, and
that 10 per cent tolerance resistors be used
for R, and R,. Also, the rectified carrier voltage developed across C, should be at least 5
volts for good limiting.
The limiter will work well on c -w telegraphy
if the amplitude of beat frequency oscillator
injection is not too high. Variable injection is
to be preferred, adjustable from the front panel.
a
D
228
THE
Receiver Fundamentals
Radio
V2
VI
LAST I. F.TUDC
I
-0.1 -µfd.
paper
mice
C_100 -44fd. mica
C4, Cs -0.01 -µfd. paper
megohm,
Rr, R2
15 watt
R,, R4- 220,000 ohms,
Vt watt
R6, R6-1 megohm,
C1
AUDIO
FT.
RADIO
C,
-50 -4µtd.
-1
4T
Rr
watt
-2- megohm potentiometer
Figure 30
NOISE LIMITER CIRCUIT, WITH ASSOCIATED A -V -C
This limiter is of the series type, and is self-adjusting to carrier strength for phone reception. For proper
operation several volts should be developed across the secondary of the last i -f transformer (IFT) under
carrier conditions.
If this feature is not provided, the b -f-o injection should be reduced to the lowest value that
will give a satisfactory beat. When this is
done, effective limiting and a good beat can
be obtained by proper adjustment of the r -f
and a -f gain controls. It is assumed, of course,
that the a.v.c. is cut out of the circuit for
c -w telegraphy reception.
Alternative
Limiter Circuit
The
more
circuit of figure 31 is
effective than that
shown in figure 30 under certain conditions and requires the addition of
only one more resistor and one more capacitor than the other circuit. Also, this circuit
involves a smaller loss in output level than
the circuit of figure 30. This circuit can be
used with equal effectiveness with a combined diode - triode or diode -pentode tube (6R7,
6SR7, 6Q7, 6SQ7 or similar diode -triodes, or
6B8, 6SF7, or similar diode -pentodes) as diode
detector and first audio stage. However, a
separate diode must be used for the noise
limiter, D,. This diode may be one -half of a
6H6, 6AL5, 7A6, etc., or it may be a triode
connected 6J5, 6C4 or similar type.
Note that the return for the volume control
must be made to the cathode of the detector
diode (and not to ground) when a dual tube is
used as combined second- detector first- audio.
This means that in the circuit shown in figure
31 a connection will exist across the points
where the "X" is shown on the diagram since
a common cathode lead is brought out of the
tube for Dr and V,. If desired, of course, a
single dual diode may be used for Dr and D,
in this circuit as well as in the circuit of figure 30. Switching the limiter in and out with
the switch S brings about no change in volume.
In any diode limiter circuit such as the ones
shown in these two figures it is important that
the mid -point of the heater potential for the
noise -limiter diode be as close to ground
potential as possible. This means that the
center -tap of the heater supply for the tubes
should be grounded wherever possible rather
than grounding one side of the heater supply
as is often done. Difficulty with hum pickup
in the limiter circuit may be encountered when
one side of the heater is grounded due to the
high values of resistance necessary in the
limiter circuit.
The circuit of figure
31 has been used with
excellent success in several home -constructed
receivers, and in the BC- 312/BC -342 and BC348 series of surplus communications receivers. It is also used in certain manufactured
receivers.
An excellent check on the operation of the
noise limiter in any communications receiver
can be obtained by listening to the Loran signals in the 160 -meter band. With the limiter
out a sharp rasping buzz will be obtained
when one of these stations is tuned in. With
the noise limiter switched into the circuit the
buzz should be greatly reduced and a low pitched hum should be heard.
The most satisafctory diode
noise limiter is the series full wave limiter, shown in figure
32. The positive noise peaks are clipped by
diode A, the clipping level of which may be
adjusted to clip at any modulation level between 25 per cent and 100 per cent. The negaative noise peaks are clipped by diode B at
The Full -Wave
Limiter
a
fixed level.
Twin Noise Squelch.
popularized by CQ magazine, is a combination of a diode noise clipper
and an audio squelch tube. The squelch cit.-
The TNS
Limiter
The
HANDBOOK
U
-H -F
Circuits
229
This circuit is of the selfadjusting type and gives less
distortion for a given degree
of modulation than the more
limiter circuits.
-470K, 14 won
R3 -100K,
watt
common
R1, R2
R.,
RS
V2
-1
megohm, VI watt
megohm potentiometer
-2mica (approx.)
c2- 0.01 -µtd. paper
R6
C1- 0.00025
C3- 0.01 -4fd.
C4- 0.01 -12fd.
paper
paper
02 -6116,
D3,
6AL5,
sections of
diode
7A6,
a
or
6S8 -GT
Figure
31
ALTERNATIVE NOISE LIMITER CIRCUIT
cuit is useful in eliminating the grinding background noise that is the residual left by the
diode clipper. In figure 33, the setting of the
470K potentiometer determines the operating
level of the squelch action and should be set
to eliminate the residual background noise.
Because of the low inherent distortion of the
TNS, it may be left in the circuit at all times.
As with other limiters, the TNS requires a
high signal level at the second detector for
maximum limiting effect.
12 -10
Special Considerations
wavelength sections of parallel conductors or
concentric transmission line are not only more
efficient but also become of practical dimensions.
Tuning
Short Lines
Tubes and tuning capacitors con netted to the open end of a transmission line provide a capacitance that makes the resonant length less than
a quarter wave -length. The amount of shortening for a specified capacitive reactance is
determined by the surge impedance of the line
in U -H -F Receiver Design
increasingly higher frequencies, it becomes progressively
more difficult to obtain a satisfactory amount of selectivity and impedance
from an ordinary coil and capacitor used as a
Transmission
At
F
STAGE
2ND DEI. -AUDIO
Line Circuits
S n. DIO
resonant circuit. On the other hand, quarter
F
T
2ND DET
AuD10
Figure 32
THE FULL -WAVE SERIES AUDIO
NOISE LIMITER
Figure 33
THE TNS AUDIO NOISE LIMITER
www.americanradiohistory.com
230
Radio
Receiver
Fundamentals
THE
RADIO
CAVITY
CAVITY
0LOOV
LINE
CONCENTRIC
LINE
CAVITY
Figure 34
COUPLING AN ANTENNA TO A
O
©
GVITY
CAVITY
GRIDS
ELECTRON
BEAM
HOLE
COAXIALRESONANT CIRCUIT
(A) shows the recommended method for coupling
a coaxial line to a coaxial resonant circuit. (B)
shows an alternative method for use with an open wire type of antenna feed line.
section. It is given by the equation for resonance:
1
2rr /C
=
Z. tan
1
77 = 3.1416, / is the frequency, C the
capacitance, Z. the surge impedance of the
line, and tan / is the tangent of the electrical
length in degrees.
The capacitive reactance of the capacitance
across the end is 1 /(277 / C) ohms. For resonance, this must equal the surge impedance of
the line times the tangent of its electrical
length (in degrees, where 90° equals a quarter wave). It will be seen that twice the capacitance will resonate a line if its surge impedance is halved; also that a given capacitance
has twice the loading effect when the frequency is doubled.
in which
It is possible to couple into
a parallel -rod line by tapping directly on one or both
rods, preferably through
blocking capacitors if any d.c. is present.
More commonly, however, a hairpin is inductively coupled at the shorting bar end, either
to the bar or to the two rods, or both. This
normally will result in a balanced load. Should
a loop unbalanced to ground be coupled in,
any resulting unbalance reflected into the rods
can be reduced with a simple Faraday screen,
made of a few parallel wires placed between
the hairpin loop and the rods. These should
be soldered at only one end and grounded.
An unbalanced tap on a coaxial resonant
Coupling Into
Lines and
Coaxial Circuits
Figure 35
METHODS OF EXCITING A RESONANT
CAVITY
circuit can be made directly on the inner conductor at the point where it is properly matched
(figure 34). For low impedances, such as a concentric line feeder, a small one -half turn loop
can be inserted through a hole in the outer
conductor of the coaxial circuit, being in effect a half of the hairpin type recommended
for coupling balanced feeders to coaxial resonant lines. The size of the loop and closeness to the inner conductor determines the
impedance matching and loading. Such loops
coupled in near the shorting disc do not alter
the tuning appreciably, if not overcoupled.
cavity is a closed resonant
made of metal. It is
known also as a rhumbatron. The
cavity, having both inductance and capacitance, supersedes coil -capacitor and capacitance- loaded transmission -line tuned circuits
at extremely high frequencies where conventional L and C components, of even the most
refined design, prove impractical because of
the tiny electrical and physical dimensions
they must have. Microwave cavities have high
Q factors and are superior to conventional
tuned circuits. They may be employed in the
manner of an absorption wavemeter or as the
tuned circuit in other r -f test instruments, and
in microwave transmitters and receivers.
Resonant cavities usually are closed on all
sides and all of their walls are made of electrical conductor. However, in some forms,
small openings are present for the purpose of
excitation. Cavities have been produced in
several shapes including the plain sphere,
Resonant
Cavities
A
chamber
www.americanradiohistory.com
HANDBOOK
TUNING
U
dimpled sphere, sphere with reentrant cones
of various sorts, cylinder, prism (including
cube), ellipsoid, ellipsoid -hyperboloid, doughnut- shape, and various reentrant types. In appearance, they resemble in their simpler forms
metal boxes or cans.
The cavity actually is a linear circuit, but
one which is superior to a conventional coaxial resonator in the s -h -f range. The cavity
resonates in much the same manner as does a
barrel or a closed room with reflecting walls.
Because electromagnetic energy, and the
associated electrostatic energy, oscillates to
and fro inside them in one mode or another,
resonant cavities resemble wave guides. The
mode of operation in a cavity is affected by
the manner in which micro-wave energy is injected. A cavity will resonate to a large number of frequencies, each being associated with
a particular mode or standing -wave pattern.
The lowest mode (lowest frequency of operation) of a cavity resonator normally is the one
used.
The resonant frequency of a cavity may be
varied, if desired, by means of movable plungers or plugs, as shown in figure 36A, or a
movable metal disc (see figure 36B). A cavity
that is too small for a given wavelength will
not oscillate.
The resonant frequencies of simple spherical, cylindrical, and cubical cavities may be
calculated simply for one particular mode.
Wavelength and cavity dimensions (in centimeters) are related by the following simple
resonance formulae:
Butterfly
231
DISC
Figure 36
TUNING METHODS FOR CYLINDRICAL
RESONANT CAVITIES
= 2.6 x
= 2.83 x
= 2.28x
radius
half of
radius
1
side
Unlike the cavity resonator, which
in its conventional form is a device
which can tune over a relatively
narrow band, the butterfly circuit is a tunable
resonator which permits coverage of a fairly
Circuit
Circuits
MOVEABLE
SLUGS
For Cylinder A,
" Cube
Ar
" Sphere A,
-H -F
Figure 37
THE BUTTERFLY RESONANT CIRCUIT
Shown at (A) is the physical appearance of the
butterfly circuit as used in the v -h -f and lower
u-h-f ronge. (B) shows an electrical representation of the circuit.
wide
u -h -f band. The butterfly circuit is very
similar to a conventional coil -variable capacitor combination, except that both inductance
and capacitance are provided by what appears
to be a variable capacitor alone. The Q of this
device is somewhat less than that of a concentric -line tuned circuit but is entirely adequate for numerous applications.
Figure 37A shows construction of a single
butterfly section. The butterfly- shaped rotor,
from which the device derives its name, turns
in relation to the unconventional stator. The
two groups of stator "fins" or sectors are in
effect joined together by a semi -circular metal
band, integral with the sectors, which provides
the circuit inductance. When the rotor is set
to fill the loop opening (the position in which
it is shown in figure 37A), the circuit inductance and capacitance are reduced to minimum.
When the rotor occupies the position indicated
by the dotted lines, the inductance and capacitance are at maximum. The tuning range of
practical butterfly circuits is in the ratio of
1.5:1 to 3.5:1.
Direct circuit connections may be made to
points A and B. If balanced operation is desired, either point C or D will provide the
electrical mid-point. Coupling may be effected
by means of a small singleturn loop placed
near point E or F. The butterfly thus permits
continuous variation of both capacitance and
inductance, as indicated by the equivalent
circuit in figure 37B, while at the same time
eliminating all pigtails and wiping contacts.
Several butterfly sections may be stacked
in parallel in the same way that variable capacitors are built up. In stacking these sections, the effect of adding inductances in parallel is to lower the total circuit inductance,
while the addition of stators and rotors raises
the total capacitance, as well as the ratio of
maximum to minimum capacitance.
www.americanradiohistory.com
232
Radio Receiver Fundamentals
Butterfly circuits have been applied specifically to oscillators for transmitters, superheterodyne receivers, and heterodyne frequency meters in the 100 - 1000 -Mc. frequency range.
The types of resonant circuits described in the previous paragraphs
have largely replaced conventional
coil -capacitor circuits in the range above 100
Mc. Tuned short lines and butterfly circuits
are used in the range from about 100 Mc. to
perhaps 3500 Mc., and above about 3500 Mc.
resonant cavities are used almost exclusively.
The resonant cavity is also quite generally
employed in the 2000 -Mc. to 3500 -Mc. range.
In a properly designed receiver, thermal
agitation in the first tuned circuit is amplified
by subsequent tubes and predominates in the
output. For good signal-to- set -noise ratio,
therefore, one must strive for a high -gain low noise r -f stage. Hiss can be held down by giving careful attention to this point. A mixer
has about 0.3 of the gain of an r -f tube of the
same type; so it is advisable to precede a
mixer by an efficient r-f stage. It is also of
some value to have good r -f selectivity before
the first detector in order to reduce noises
produced by beating noise at one frequency
against noise at another, to produce noise at
the intermediate frequency in a superheterodyne.
The frequency limit of a tube is reached
when the shortest possible external connections are used as the tuned circuit, except for
abnormal types of oscillation. Wires or sizeable components are often best considered as
sections of transmission lines rather than as
simple resistances, capacitances, or inductReceiver
Circuits
ances.
as small triodes and pentodes will
operate normally, they are generally preferred
as v-h -f tubes over other receiving methods
that have been devised. However, the input
capacitance, input conductance, and transit
time of these tubes limit the upper frequency
at which they may be operated. The input resistance, which drops to a low value at very
short wave -lengths, limits the stage gain and
broadens the tuning.
So long
The first tube in a v -h -f receiver is
most important in raising the signal
above the noise generated in successive stages, for which reason small v -h-f types
are definitely preferred.
Tubes employing the conventional grid-controlled and diode rectifier principles have been
modernized, through various expedients, for
operation at frequencies as high, in some new
types, as 4000 Mc. Beyond that frequency,
electron transit time becomes the limiting facV -H -F
Tubes
THE
RADIO
tor and new principles must be enlisted. In
general, the improvements embodied in existing tubes have consisted of (1) reducing electrode spacing to cut down electron transit
time, (2) reducing electrode areas to decrease
interelectrode capacitances, and (3) shortening of electrode leads either by mounting the
electrode assembly close to the tube base or
by bringing the leads out directly through the
glass envelope at nearby points. Through reduction of lead inductance and interelectrode
capacitances, input and output resonant frequencies due to tube construction have bee':
increased substantially.
Tubes embracing one or more of the features just outlined include the later !octal
types, high- frequency acorns, button -base
types, and the lighthouse types. Type 6J4
button-base triode will reach 500 Mc. Type
6F4 acorn triode is recommended for use up to
1200 Mc. Type 1A3 button -base diode has a
resonant frequency of 1000 Mc., while type
9005 acorn diode resonates at 1500 Mc. Lighthouse type 2C40 can be used at frequencies
up to 3500 Mc. as an oscillator.
More than two de c a d e s have
passed since the crystal (mineral)
rectifier enjoyed widespread use
in radio receivers. Low -priced tubes completely supplanted the fragile and relatively insensitive crystal detector, although it did continue for a few years as a simple meter rectifier in absorption wavemeters after its demise
as a receiver component.
Today, the crystal detector is of new importance in microwave communication. It is
being employed as a detector and as a mixer
in receivers and test instruments used at ex-
Crystal
Rectifiers
tremely high radio frequencies. At some of
the frequencies employed in microwave operations, the crystal rectifier is the only satisfactory detector or mixer. The chief advantages of the crystal rectifier are very low capacitance, relative freedom from transit -time
difficulties, and its two -terminal nature. No
batteries or a -c power supply are required for
its operation.
The crystal detector consists essentially of
a small piece of silicon or germanium mounted in a base of low- melting -point alloy and
contacted by means of a thin, springy feeler
wire known as the cat whisker. This arrangement is shown in figure 38A.
The complex physics of crystal rectification
is beyond the scope of this discussion. It is
sufficient to state that current flows from several hundred to several thousand times more
readily in one direction through the contact of
cat whisker and crystal than in the opposite
direction. Consequently, an alternating current
(including one of microwave frequency) will
www.americanradiohistory.com
HANDBOOK
SYMBOL
f
Receiver Adjustment
\'i
BRASS BASE CONNECTOR
-CERAMIC SLEEVE
BRASS CAP
BRASS CONNECTOR PIN
CRYSTAL DIODE
small silicon crystal is attached to
the base connector and o fine "cat whisker" wire is set to the most sensitive spot on the crystal. After adjustment the ceramic shell is filled with
compound to hold the contact wire in
position. Crystals of this type are used
to over 30,000 mc.
A
rectified by the crystal detector. The load,
through which the rectified currents flow, may
be connected in series or shunt with the crystal, although the former connection is most
generally employed.
The basic arrangement of a modern fixed
crystal detector developed during World War II
for microwave work, particularly radar, is
shown in figure 38B. Once the cat whisker of
this unit is set at the factory to the most sensitive spot on the surface of the silicon crystal and its pressure is adjusted, a filler compound is injected through the filling hole to
hold the cat whisker permanently in position.
be
Receiver Adjustment
simple regenerative receiver requires
adjustment other than that necessary
to insure correct tuning and smooth regeneration over some desired range. Receivers of
the tuned radio- frequency type and superheterodynes require precise alignment to obtain the
highest possible degree of selectivity and
A
little
sensitivity.
Good results can be obtained from a receiver only when it is properly aligned and adjusted. The most practical technique for mak-
ing these adjustments is given below.
A very small number
ments
Alignment procedure in a multistage t -r -f receiver is exactly the same as aligning a
single stage. If the detector is regenerative,
each preceding stage is successively aligned
while keeping the detector circuit tuned to the
test signal, the latter being a station signal
or one locally generated by a test oscillator
loosely coupled to the antenna lead. During
these adjustments, the r -f amplifier gain control is adjusted for maximum sensitivity, assuming that the r-f amplifier is stable and
does not oscillate. Often a sensitive receiver
can be roughly aligned by tuning for maximum
noise pickup.
Alignment
1N23 MICROWAVE -TYPE
Instruments
receiver output when tuning to a modulated
signal. If the signal is a steady tone, such as
from a test oscillator, the output meter will
indicate the value of the detected signal. In
this manner, alignment results may be visually
noted on the meter.
T -R -F Receiver
Figure 38
12 -11
233
of instru-
will suffice to check
and
align a communications receiver, the most important of these testing units being a modulated oscillator and a d -c and a -c voltmeter.
The meters are essential in checking the voltage applied at each circuit point from the power supply. If the a -c voltmeter is of the oxide rectifier type, it can be used, in addition, as
an output meter when connected across the
Aligning a superhet is a detailed task requiring a great
amount of care and patience.
It should never be undertaken without a thorough understanding of the involved job to be
done and then only when there is abundant
Superheterodyne
Alignment
time to devote to the operation. There are no
short cuts; every circuit must be adjusted individually and accurately if the receiver is to
give peak performance. The precision of each
adjustment is dependent upon the accuracy
with which the preceding one was made.
Superhet alignment requires (1) a good signal generator (modulated oscillator) covering
the radio and intermediate frequencies and
equipped with an attenuator; (2) the necessary
socket wrenches, screwdrivers, or "neutralizing tools" to adjust the various i -f and r -f
trimmer capacitors; and (3) some convenient
type of tuning indicator, such as a copper oxide or electronic voltmeter.
Throughout the alignment process, unless
specifically stated otherwise, the r -f gain control must be set for maximum output, the beat
oscillator switched off, and the a.v.c. turned
off or shorted out. When the signal output of
the receiver is excessive, either the attenuator
or the a -f gain control may be turned down, but
never the r -f gain control.
After the receiver has been
given a rigid electrical and
mechanical inspection, and any faults which
may have been found in wiring or the selection and assembly of parts corrected, the i -f
amplifier may be aligned as the first step in
the checking operations.
Vl ith the signal generator set to give a modulated signal on the frequency at which the i -f
I
-F Alignment
www.americanradiohistory.com
234
amplifier is to operate, clip the "hot" output
lead from the generator to the last i -f stage
through a small fixed capacitor to the control
grid. Adjust both trimmer capacitor's in the
last i -f transformer (the one between the last
i -f amplifier and the second detector) to resonance as indicated by maximum deflection of
the output meter.
Each i -f stage is adjusted in the same manner, moving the hot lead, stage by stage, back
toward the front end of the receiver and backing off the attenuator as the signal strength
increases in each new position. The last adjustment will be made to the first i -f transformer, with the hot signal generator lead connected to the control grid of the mixer. Occasionally it is necessary to disconnect the
mixer grid lead from the coil, grounding it
through a 1,000- or 5,000 -ohm resistor, and
coupling the signal generator through a small
capacitor to the grid.
When the last i -f adjustment has been completed, it is good practice to go back through
the i -f channel, re- peaking all of the transformers. It is imperative that this recheck be
made in sets which do not include a crystal
filter, and where the simple alignment of the
i -f amplifier to the generator is final.
I
RADIO
THE
Radio Receiver Fundamentals
-F SIGNAL IN
ALONE
t
w
`
F
PLUS
O
MULTIPLIER
f
W
f
55
CC
FREQUENCY
Figure
39
THE Q- MULTIPLIER
The loss resistance of a high -Q
neutralized by regeneration
is
circuit
in a
feedback amplifier. A highly
selective passband is produced which
circuit of the
is coupled to the i -f
receiver.
simple
-F with
Crystal Filter
There are several ways of align ing an i -f channel which contains a crystal -filter circuit.
However, the following method is one which
has been found to give satisfactory results in
every case: An unmodulated signal generator
capable of tuning to the frequency of the filter
crystal in the receiver is coupled to the grid of
the stage which precedes the crystal filter in
the receiver. Then, with the crystal filter
switched in, the signal generator is tuned
slowly to find the frequency where the crystal
peaks. The receiver "S" meter may be used
as the indicator, and the sound heard from the
loudspeaker will be of assistance in finding
the point. When the frequency at which the
crystal peaks has been found, all the i -f transformers in the receiver should be touched up
to peak at that frequency.
I
Adjusting the beat oscillator on a receiver that has
no front panel adjustment is relatively simple.
It is only necessary to tune the receiver to
resonance with any signal, as indicated by
the tuning indicator, and then turn on the b.f.o.
B -F -Q
Adjustment
and set its trimmer (or trimmers) to produce
the desired beat note. Setting the beat oscil-
lator in this way will result in the beat note
being stronger on one "side" of the signal
than on the other, which is what is desired
for c -w reception. The b.f.o. should not be set
to zero beat when the receiver is tuned to
resonance with the signal, as this will cause
an equally strong beat to be obtained on both
sides of resonance.
Alignment of the front end of a
receiver is a
relatively simple process, consisting of first getting the oscillator to cover
the desired frequency range and then of peaking the various r -f circuits for maximum gain.
However, if the frequency range covered by
the receiver is very wide a fair amount of cut
and try will be required to obtain satisfactory
tracking between the r-f circuits and the oscillator. Manufactured communications receivers
should always be tuned in accordance with
the instructions given in the maintenance manual for the receiver.
Front -End
Alignment
12 -12
home -constructed
Receiving Accessories
The selectivity of a receiver
may be increased by raising
the Q of the tuned circuits of the i -f strip. A
simple way to accomplish this is to add a controlled amount of positive feedback to a tuned
circuit, thus increasing its Q. This is done in
the 0- multiplier, whose basic circuit is shown
in figure 39. The circuit L -C1 -C2 is tuned to
The Q- Multiplier
www.americanradiohistory.com
Receiving Accessories 235
HANDßCOK
I.F SIGNAL IN
I
-F SIGNAL OJT
.005
6
_
L2
Lt
.005
NULL
O
12 10X7
TO PLATE TERMINAL
OF FIRST I-F TUBE TNRU
2. OF COAXIAL LINE
PEAR
MULTIPLIER "NULL"
SELECTIfIrY
2 12Ax7
CONTROLS
3
32 MEG
1.SK
10K
I
LI. GRAY6'URNE
455 KC
FREQUENCY
Figure
MULTIPLIER NULL CIRCUIT
1-
n
T
6.3 V.
(0.6-6.OAIN)
intermediate frequency, and the loss resistance of the circuit is neutralized by the
positive feedback circuit composed of C3 and
the vacuum tube. Too great a degree of positive
feedback will cause the circuit to break into
LI is required to tune out the
reactance of the coaxial line. It is adjusted for maximum signal response.
LI may be omitted if the Q- multiplier
is connected to the receiver with o
short length of wire, and the i -f transformer within the receiver is retuned.
Coil
the
oscillation.
At the resonant frequency, the impedance of
the tuned circuit is very high, and when shunted
across an i -f stage will have little effect upon
the signal. At frequencies removed from resonance, the impedance of the circuit is low, resulting in high attenuation of the i -f signal.
The resonant frequency of the Q- multiplier may
by varied by changing the value of one of the
components in the tuned circuit.
The Q- multiplier may also be used to "null"
a signal by employing negative feedback to
control the plate resistance of an auxiliary
amplifier stage as shown in figure 40. Since the
grid- cathode phase shift through the Q- multiplier
is zero, the plate resistance of a second tube
may be readily controlled by placing it across
the Q- multiplier. At resonance, the high negative feedback drops the plate resistance of
V2, shunting the i -f circuit. Off resonance, the
feedback is reduced and the plate resistance
of V2 rises, reducing the amount of signal attenuation in the i -f strip. A circuit combining
both the "peak" and "null" features is shown
in figure 41.
A version of the common
mixer or converter stage
41
SCHEMATIC OF A 455KC
0- MULTIPLIER
The addition of a second triode permits
the 0- Multiplier to be used for nulling
out an unwanted hetrodyne.
The Product Detector
V.
L22GRAY6URNE "LOOPSTICK" COIL
Figure 40
Q-
+200-300
B
V6 CNOKE
A.
may be used as a second detector in a receiver
in place of the usual diode detector. The diode
is an envelope detector (section 12 -1) and develops a d -c output voltage from a single r -f
signal, and audio "beats" from two or more
input signals. A product detector (figure 42)
requires that a local carrier voltage be present
in order to produce an audio output signal.
R
PRODUCT
DETECTOR
-F SIGNAL
LOCAL
OSCILLATOR
Figure 42
THE PRODUCT
DETECTOR
Audio output signal is
when
developed
only
local oscillator is on.
www.americanradiohistory.com
HAUDIO OUTPUT
Radio Receiver Fundamentals
236
Va
V,
I
-F SIGNAL
VS
12AU7
12AU7
I-F
0,
r+AU01O OUT
4711
SIC.
SO
SEAT OSC.
SIGNAL
Figure
Figure 43
PENTAGRID MIXER
USED AS PRODUCT
DETECTOR
PRODUCT DETECTOR
VI and V2 act as cathode followers, delivering sideband signal
and local oscillator signal to
grounded grid triode mixer (V3).
Such a detector is useful for single sideband
work, since the inter -modulation distortion is
extremely low.
A pentagrid product detector is shown in
figure 43. The incoming signal is applied to
grid 3 of the mixer tube, and the local oscillator
is injected on grid 1. Grid bias is adjusted for
operation over the linear portion of the tube
characteristic curve. When grid 1 injection
is removed, the audio output from an unmodulated signal applied to grid 3 should be reduced
approximately 30 to 40 db below normal detection level. When the frequency of the local
oscillator is synchronized with the incoming
carrier, amplitude modulated signals may be
received by exalted carrier reception, wherein
the local carrier substitutes for the transmitted
carrier of the a -m signal.
Three triodes may be used as a product
detector (figure 44). Triodes V1 and V2 act
as cathode followers, delivering the sideband
signal and the local oscillator signal to a
grounded grid triode (V3) which functions as
the mixer stage. A third version of the product
detector is illustrated in figure 45. A twin
triode tube is used. Section V1 functions as
a cathode follower amplifier. Section V2 is a
.
?tl
ì
II14611
i.h°1
..
IIIIII
44
TRIPLE -TRIODE
Figure 45
DOUBLE -TRIODE
PRODUCT DETECTOR
"plate"
detector, the cathode of which is
with the cathode follower amplifier.
The local oscillator signal is injected into the
grid circuit of tube V2.
common
Figure 46
EXPLODED VIEW OF COLLINS
MECHANICAL FILTER
.1. 'If#I!IIItIYIIiIIIIvIIIII'I+II
vIIl°1i
www.americanradiohistory.com
1
1
CHAPTER THIRTEEN
Generation of
Radio Frequency Energy
A radio communication or broadcast transmitter consists of a source of radio frequency
power, or carrier; a system for modulating the
carrier whereby voice or telegraph keying or
other modulation is superimposed upon it; and
an antenna system, including feed line, for
radiating the intelligence- carrying radio frequency power. The power supply employed to
convert primary power to the various voltages
required by the r -f and modulator portions of
the transmitter may also be considered part
of the transmitter.
Voice modulation usually is accomplished
by varying either the amplitude or the frequency of the radio frequency carrier in accordance
with the components of intelligence to be
transmitted.
Radiotelegraph modulation (keying) normally is accomplished either by interrupting,
shifting the frequency of, or superimposing an
audio tone on the radio -frequency carrier in
accordance with the dots and dashes to be
transmitted.
The complexity of the radio- frequency generating portion of the transmitter is dependent
upon the power, order of stability, and frequency desired. An oscillator feeding an antenna
directly is the simplest form of radio- frequency
generator. A modern high -frequency transmitter,
on the other hand, is a very complex generator.
Such an equipment usually comprises a very
stable crystal -controlled or self-controlled
oscillator to stabilize the output frequency, a
series of frequency multipliers, one or more
amplifier stages to increase the power up to
the level which is desired for feeding the antenna system, and a filter system for keeping
the harmonic energy generated in the transmitter from being fed to the antenna system.
13
-1
Controlled
Oscillators
Self-
In Chapter Four, it was explained that the
amplifying properties of a tube having three
or more elements give it the ability to generate an alternating current of a frequency determined by the components associated with
it. A vacuum tube operated in such a circuit
is called an oscillator, and its function is
essentially to convert direct current into radio frequency alternating current of a predetermined frequency.
Oscillators for controlling the frequency of
conventional radio transmitters can be divided
into two general classes: self-controlled and
crystal- controlled.
There are a great many types of self-controlled oscillators, each of which is best suited
237
www.americanradiohistory.com
Generation of
238
OA SHUNT -FED HARTLEY
R
-F
Energy
THE
OB SHUNT -FED COLPITTS
R
©
RADIO
TUNED PLATE TUNED GRID
R
L+
250
Li
250
GRID
COIL
OD TUNED -PLATE UNTUNED GRID
EO
ELECTRON COUPLED
HO
CLAPP ELECTRON COUPLED
FO
COLPITTS ELECTRON COUPLED
L
OG CLAPP
Figure
1
COMMON TYPES OF SELF -EXCITED OSCILLATORS
Fixed capacitor values are typical, but will vary somewhat with the application.
In the Clapp oscillator circuits (G) and (H), capacitors Cr and C2 should have
reactance of 50 to 100 ohms at the operating frequency of the oscillator. Tuning ofa
these two oscillators is accomplished by capacitor C. In the circuits of (E), (F), and
(H), tuning of the tank circuit in the plate of the oscillator tube will have relatively
small effect on the frequency of oscillation. The plate tank circuit also may, if desired, be tuned to a harmonic of the oscillation frequency, or a broadly resonant
circuit may be used in this circuit position.
to a
particular application. They can further
be subdivided into the
classifications
of: neg-
ative -grid oscillators, electron -orbit oscillators, negative -resistance oscillators, velocity
modulation oscillators, and magnetron oscillators.
negative -grid oscillator is
a vacuum-tube amplifier with a sufficient portion of the output energy coupled back into the
input circuit to sustain oscillation. The conNegative -Grid
A
Oscillators
essentially
trol grid is biased negatively with respect to
the cathode. Common types of negative -grid
oscillators are diagrammed in figure 1.
Illustrated in figure 1 (A) is the
oscillator circuit which finds the
most general application at the present time;
this circuit is commonly called the Hartley.
The operation of this oscillator will be described as an index to the operation of all
negative -grid oscillators; the only real differThe Hartley
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Oscillators
HANDBOOK
ence between the various circuits is the manner in which energy for excitation is coupled
from the plate to the grid circuit.
When plate voltage is applied to the Hartley
oscillator shown at (A), the sudden flow of
plate current accompanying the application of
plate voltage will cause an electro- magnetic
field to be set up in the vicinity of the coil.
The building -up of this field will cause a potential drdp to appear from turn-to -turn along
the coil. Due to the inductive coupling between the portion of the coil in which the plate
current is flowing and the grid portion, a potential will be induced in the grid portion.
Since the cathode tap is between the grid
and plate ends of the coil, the induced grid
voltage acts in such a manner as to increase
further the plate current to the tube. This action will continue for a short period of time
determined by the inductance and capacitance
of the tuned circuit, until the flywheel effect
of the tuned circuit causes this action to come
to a maximum and then to reverse itself. The
plate current then decreases, the magnetic
field around the coil also decreasing, until a
minimum is reached, when the action starts
again in the original direction and at a greater
amplitude than before. The amplitude of these
oscillations, the frequency of which is determined by the coil -capacitor circuit, will increase in a very short period of time to a limit
determined by the plate voltage of the oscil-
lator tube.
(8) shows a version of
oscillator. It can
be seen that this is essentially the same circuit as the Hartley except that the ratio of a
The Colpitts
Figure
1
the Colpitts
pair of capacitances in series determines the
effective cathode tap, instead of actually using a tap on the tank coil. Also, the net capacitance of these two capacitors comprises
the tank capacitance of the tuned circuit. This
oscillator circuit is somewhat less susceptible to parasitic (spurious) oscillations than
the Hartley.
For best operation of the Hartley and Colpitts oscillators, the voltage from grid to cathode, determined by the tap on the coil or the
setting of the two capacitors, normally should
be from 1/3 to 1/5 that appearing between
plate and cathode.
The tuned -plate tuned-grid oscillator illustrated at (C) has
a tank circuit in both the plate and grid circuits. The feedback of energy from the plate
to the grid circuits is accomplished by the
The T.P.T.G.
plate -to-grid inter -electrode capacitance within the tube. The necessary phase reversal in
feedback voltage is provided by tuning the
grid tank capacitor to the low side of the de-
239
sired frequency and the plate capacitor to the
high side. A broadly resonant coil may be substituted for the grid tank to form the T.N. T.
oscillator shown at (D).
Electron -Coupled
In any of the
oscillator circuits just described it is
possible to take energy from
the oscillator circuit by coupling an external
load to the tank circuit. Since the tank circuit
determines the frequency of oscillation of the
tube, any variations in the conditions of the
external circuit will be coupled back into the
frequency determining portion of the oscillator.
These variations will result in frequency instability.
The frequency determining portion of an
oscillator may be coupled to the load circuit
only by an electron stream, as illustrated in
(E) and (F) of figure 1. When it is considered
that the screen of the tube acts as the plate
to the oscillator circuit, the plate merely actOscillators
ing as a coupler to the load, then the similarity between the cathode -grid- screen circuit
of these oscillators and the cathode -grid -plate
circuits of the corresponding prototype can be
seen.
The electron- coupled oscillator has good
stability with respect to load and voltage variation. Load variations have a relatively small
effect on the frequency, since the only coupling between the oscillating circuit and the
load is through the electron stream flowing
through the other elements to the plate. The
plate is electrostatically shielded from the
oscillating portion by the bypassed screen.
The stability of the e.c.o. with respect to
variations in supply voltages is explained as
follows: The frequency will shift in one direction with an increase in screen voltage, while
an increase in plate voltage will cause it to
shift in the other direction. By a proper proportioning of the resistors that comprise the
voltage divider supplying screen voltage, it is
possible to make the frequency of the oscillator substantially independent of supply voltage variations.
The Clapp
Oscillator
relatively new type of oscillator
circuit which is capable of giving
excellent frequency stability is
A
illustrated in figure 1G. Comparison between
the more standard circuits of figure IA through
IF and the Clapp oscillator circuits of figures
1G and 1H will immediately show one marked
difference: the tuned circuit which controls
the operating frequency in the Clapp oscillator
is series resonant, while in all the more standard oscillator circuits the frequency controlling circuit is parallel resonant. Also, the
capacitors C, and C, are relatively large in
terms of the usual values for a Colpitts oscil-
240
Generation of
R
-F
THE
Energy
lator. In fact, the value of capacitors C, and
C, will be in the vicinity of 0.001 µfd. to
0.0025 µfd. for an oscillator which is to be
operated in the 1.8 -Mc. band.
The Clapp oscillator operates in the following manner: at the resonant frequency of the
oscillator tuned circuit (L, C) the impedance
of this circuit is at minimum (since it operates in series resonance) and maximum current flows through it. Note however, that C,
and C, also are included within the current
path for the series resonant circuit, so that at
the frequency of resonance an appreciable
voltage drop appears across these capacitors.
The voltage drop appearing across C, is applied to the grid of the oscillator tube as excitation, while the amplified output of the
oscillator tube appears across C, as the driving power to keep the circuit in oscillation.
Capacitors C, and C, should be made as
large in value as possible, while still permitting the circuit to oscillate over the full tuning range of C. The larger these capacitors
are made, the smaller will be the coupling between the oscillating circuit and the tube, and
consequently the better will be oscillator stability with respect to tube variations. High Gm
tubes such as the 6AC7, 6ÁG7, and 6CB6 will
permit the use of larger values of capacitance
at C, and C, than will more conventional tubes
such as the 6SJ7, 6V6, and such types. In general it may be said that the reactance of capacitors C, and C, should be on the order of
40 to 120 ohms at the operating frequency of
the oscillator -with the lower values of reactance going with high -Gm tubes and the
higher values being necessary to permit oscillation with tubes having Gm in the range of
2000 micromhos such as the 6SJ7.
It will be found that the Clapp oscillator
will have a tendency to vary in power output
over the frequency range of tuning capacitor
C. The output will be greatest where C is at
its largest setting, and will tend to fall off
with C at minimum capacitance. In fact, if
capacitors C, and C, have too large a value
the circuit will stop oscillation near the minimum capacitance setting of C. Hence it will
be necessary to use a slightly smaller value
of capacitance at C, and C, (to provide an increase in the capacitive reactance at this
point), or else the frequency range of the oscillator must be restricted by paralleling a fixed
capacitor across C so that its effective capacitance at minimum setting will be increased to
a value which will sustain oscillation.
In the triode Clapp oscillator, such as shown
at figure 1G, output voltage for excitation of
an amplifier, doubler, or isolation stage normally is taken from the cathode of the oscillator tube by capacitive coupling to the grid
of the next tube. However, where greater iso-
RADIO
lation of succeeding stages from the oscillating circuit is desired, the electron- coupled
Clapp oscillator diagrammed in figure 1H may
be used. Output then may be taken from the
plate circuit of the tube by capacitive coupling
with either a tuned circuit, as shown, or with
an r -f choke or a broadly resonant circuit in
the plate return. Alternatively, energy may be
coupled from the output circuit L,-C, by link
coupling. The considerations with regard to
and the grid tuned circuit are the same
as for the triode oscillator arrangement of
CC
figure 1G.
Negative- resistance oscillators often are used when
unusually high frequency
stability is desired, as in a frequency meter.
The dynatron of a few years ago and the newer
transitron are examples of oscillator circuits
which make use of the negative resistance
characteristic between different elements in
some multi -grid tubes.
In the dynatron, the negative resistance is a
consequence of secondary emission of electrons from the plate of a tetrode tube. By a
proper proportioning of the electrode voltage,
an increase in screen voltage will cause a
decrease in screen current, since the increased
screen voltage will cause the screen to attract
a larger number of the secondary electrons
emitted by the plate. Since the net screen current flowing from the screen supply will be
decreased by an increase in screen voltage,
it is said that the screen circuit presents a
negative resistance.
If any type of tuned circuit, or even a resistance- capacitance circuit, is connected in
series with the screen, the arrangement will
oscillate -provided, of course, that the external
circuit impedance is greater than the negative
resistance. A negative resistance effect similar to the dynatron is obtained in the transitron
circuit, which uses a pentode with the suppressor coupled to the screen. The negative resistance in this case is obtained from a combination of secondary emission and inter -electrode coupling, and is considerably more stable
than that obtained from uncontrolled secondary
emission alone in the dynatron. A representative transitron oscillator circuit is shown in
figure 2.
The chief distinction between a conventional negative grid oscillator and a negative
resistance oscillator is that in the former the
tank circuit must act as a phase inverter in
order to permit the amplification of the tube
to act as a negative resistance, while in the
latter the tube acts as its own phase inverter.
Thus a negative resistance oscillator requires
only an untapped coil and a single capacitor
Negative Resistance Oscillators
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HANDBOOK
Oscillators
6SK7
TWO -TERMINAL
241
Figure 2
OSCILLATOR CIRCUITS
Both circuits may be used for an audio
oscillator or for frequencies into the
v -h -f range simply by placing a tank circuit tuned to the proper frequency where
indicated on the drawing. Recommended
values for the components are given below for both oscillators.
O
6SN7
TRANSITRON OSCILLATOR
OR
6J6
TRANSITION OSCILLATOR
0.01 -µfd. mica for r.f. 10 -µfd. elect. for c.f.
C2- 0.00005 -µfd. mica for r.f. 0.1 -4fd. paper for
a.f.
C3- 0.003-µfd. mico for r.f. 03-4fd. paper for
a.f.
C4- 001 -µfd. mica for r.f. 8 -4fd. elect. for a.f.
R3-220K S5 -watt carbon
R2 1800 ohms 55-watt carbon
R3 -22K 2 -watt carbon
R4 -22K 2 -watt carbon
C,-
CATHODE -COUPLED OSCILLATOR
C3- 0.00005 -µfd. mica for r.f. 0.1 -µfd. paper
for audio
C2- 0.003 -µfd.
mica for
audio
O
R1-47K
CATHODE COUPLED OSCILLATOR
as the frequency determining tank circuit, and
is classed as a two terminal oscillator. In fact,
the time constant of an R/C circuit may be
used as the frequency determining element and
such an oscillator is rather widely used as a
tunable audio frequency oscillator.
The Franklin oscillator makes
use of two cascaded tubes to
obtain the negative- resistance
effect (figure 3). The tubes may be either a
pair of triodes, tetrodes, or pentodes, a dual
The Fronklin
Oscillator
triode, or a combination of a triode and a multi grid tube. The chief advantage of this oscillator circuit is that the frequency determining
tank only has two terminals, and one side of
the circuit is grounded.
The second tube acts as a phase inverter to
give an effect similar to that obtained with the
dynatron or transitron, except that the effective
transconductance is much higher. If the tuned
circuit is omitted or is replaced by a resistor,
the circuit becomes a relaxation oscillator or
a multivibrator.
The Clapp oscillator has proved
to be inherently the most stable
of all the oscillator circuits discussed above, since minimum coupling between the oscillator tube and its associated
tuned circuit is possible. However, this inOscillator
Stability
R2
-1K
rd.
8 -pfd.
lect.
for
-watt carbon
-watt carbon
S5
1
herently good stability is with respect to tube
variations; instability of the tuned circuit with
respect to vibration or temperature will of
course have as much effect on the frequency of
oscillation as with any other type of oscillator
circuit. Solid mechanical construction of the
components of the oscillating circuit, along
with a small negative -coefficient compensating
capacitor included as an element of the tuned
circuit, usually will afford an adequate degree
of oscillator stability.
Figure 3
THE FRANKLIN OSCILLATOR CIRCUIT
A separate phase inverter tube is used in
this oscillator to feed a portion of the output
back to the input in the proper phase to sustain oscillation. The values of Cr and C2
should be as small as will permit oscillations
to be sustained over the desired frequency
range.
242
Generation of
R
-F
used to control the frequency of a transmitter in
which there are stringent
limitations on frequency tolerance, several precautions are taken to ensure that a variable
frequency oscillator will stay on frequency.
The oscillator is fed from a voltage regulated
power supply, uses a well designed and temperature compensated tank circuit, is of rugged
mechanical construction to avoid the effects
of shock and vibration, is protected against
excessive changes in ambient room temperature, and is isolated from feedback or stray
coupling from other portions of the transmitter
by shielding, filtering of voltage supply leads,
and incorporation of one or more buffer- amplifier stages. In a high power transmitter a small
amount of stray coupling from the final amplifier to the oscillator can produce appreciable
degradation of the oscillator stability if both
are on the same frequency. Therefore, the oscillator usually is operated on a subharmonic
of the transmitter output frequency, with one
or more frequency multipliers between the oscillator and final amplifier.
V. F.O.
Transmit -
When
ter Controls
13 -2
Quartz Crystal
Oscillators
Quartz is a naturally occuring crystal having a structure such that when plates are cut
relationships to the crystallographic axes, these plates will show the
piezoelectric effect -the plates will be deformed in the influence of an electric field,
and, conversely, when such a plate is compressed or deformed in any way a potential
difference will appear upon its opposite sides.
The crystal has mechanical resonance, and
will vibrate at a very high frequency because
of its stiffness, the natural period of vibration
depending upon the dimensions, the method of
electrical excitation, and crystallographic
orientation. Because of the piezoelectric properties, it is possible to cut a quartz plate
which, when provided with suitable electrodes,
will have the characteristics of a series resonant circuit with a very high L/C ratio and
very high Q. The Q is several times as high
as can be obtained with an inductor -capacitor
combination in conventional physical sizes.
The equivalent electrical circuit is shown in
figure 4A, the resistance component simply
being an acknowledgment of the fact that the
Q, while high, does not have an infinite value.
The shunt capacitance of the electrodes and
associated wiring (crystal holder and socket,
plus circuit wiring) is represented by the dotted portion of figure 4B. In a high frequency
in certain definite
THE
Energy
CI
RADIO
L
(SMALL)
Lt
(LARGE)
4: C2
I
(sraAr
SHUNT)
RI
(SMALL)
-J
Figure
4
EQUIVALENT ELECTRICAL CIRCUIT OF
QUARTZ PLATE IN A HOLDER
At (A) is shown the equivalent series -resonant circuit of the crystal itself, at (B) is
shown how the shunt capacitance of the
holder electrodes and associated wiring affects the circuit to the combination circuit
of (C) which exhibits both series resonance
and parallel resonance (anti -resonance), the
separation in frequency between the two
modes being very small and determined by
the ratio of C1 to C,.
crystal this will be considerably greater than
the capacitance component of an equivalent
series L/C circuit, and unless the shunt capacitance is balanced out in a bridge circuit,
the crystal will exhibit both resonant (series
resonant) and anti- resonant (parallel resonant)
frequencies, the latter being slightly higher
than the series resonant frequency and ap-
proaching it as C, is increased.
The series resonance characteristic is employed in crystal filter circuits in receivers
and also in certain oscillator circuits wherein
the crystal is used as a selective feedback
element in such a manner that the phase of the
feedback is correct and the amplitude adequate only at or very close to the series resonant frequency of the crystal.
While quartz, tourmaline, Rochelle salts,
ADP, and EDT crystals all exhibit the piezoelectric effect, quartz is the material widely
employed for frequency control.
As the cutting and grinding of quartz plates
has progressed to a high state of development
and these plates may be purchased at prices
which discourage the cutting and grinding by
simple hand methods for one's own use, the
procedure will be only lightly touched upon
here.
The crystal blank is cut from the raw quartz
at a predetermined orientation with respect to
the optical and electrical axes, the orientation
determining the activity, temperature coefficient, thickness coefficient, and other characteristics. Various orientations or "cuts" having useful characteristics are illustrated in
figure 5.
www.americanradiohistory.com
Crystal Oscillators
HANDBOOK
neo
243
T[.neror[ co rmus..T
eD fllT[eS
fe[0U[CT A7.11
OSCILLATORS
MK..
LOW feCWCMCT
AT
ST
Figure
5
CT
34.
¡
.N`
T
CT.DT,[T,fT
SSiS'
IT -ST.
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er
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URUC
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,
ORIENTATION OF THE
y
COMMON CRYSTAL CUTS
Tóir[eTUet
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ó
veD.tl
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a` OSCILLATORS
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a
r
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o
nA
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.r.D filTCeS
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cDC
The crystal blank is then rough -ground almost to frequency, the frequency increasing
in inverse ratio to the oscillating dimension
(usually the thickness). It is then finished to
exact frequency either by careful lapping, by
etching, or plating. The latter process consists of finishing it to a frequency slightly
higher than that desired and then silver plating
the electrodes right on the crystal, the frequency decreasing as the deposit of silver is
increased. If the crystal is not etched, it must
be carefully scrubbed and "baked" several
times to stabilize it, or otherwise the frequency and activity of the crystal will change with
time. Irradiation by X -rays recently has been
used in crystal finishing.
Unplated crystals usually are mounted in
pressure holders, in which two electrodes are
held against the crystal faces under slight
pressure. Unplated crystals also are sometimes mounted in an air -gap holder, in which
there is a very small gap between the crystal
and one or both electrodes. By making this
gap variable, the frequency of the crystal may
be altered over narrow limits (about 0.3% for
certain types).
The temperature coefficient of frequency for
various crystal cuts of the " -T" rotated family is indicated in figure 5. These angles are
typical, but crystals of a certain cut will vary
slightly. By controlling the orientation and dimensioning, the turning point (point of zero
temperature coefficient) for a BT cut plate may
be made either lower or higher than the 75 degrees shown. Also, by careful control of axes
and dimensions, it is possible to get AT cut
crystals with a very flat temperature- frequency
characteristic.
S,
CCnTUT
.T
f
..[ee
, .
_
lTtes
rr
-
[.
The first quartz plates used were either Y
cut or X cut. The former had a very high temperature coefficient which was discontinuous,
causing the frequency to jump at certain critical temperatures. The X cut had a moderately
bad coefficient, but it was more continuous,
and by keeping the crystal in a temperature
controlled oven, a high order of stability could
be obtained. However, the X cut crystal was
considerably less active than the Y cut, especially in the case of poorly ground plates.
For frequencies between 500 kc. and about
6 Mc., the AT cut crystal now is the most
widely used. It is active, can be made free
from spurious responses, and has an excellent
temperature characteristic. However, above
about 6 Mc. it becomes quite thin, and a difficult production job. Between 6 Mc. and about
12 Mc., the BT cut plate is widely used. It
also works well between 500 kc. and 6 Mc.,
but the AT cut is more desirable when a high
order of stability is desired and no crystal
oven is employed.
For low frequency operation on the order of
100 kc., such as is required in a frequency
standard, the GT cut crystal is recommended,
though CT and DT cuts also are widely used
for applications between 50 and 500 kc. The
CT, DT, and GT cut plates are known as contour cuts, as these plates oscillate along the
long dimension of the plate or bar, and are
much smaller physically than would be the
case for a regular AT or BT cut crystal for
the same frequency.
Crystal Holders
Crystals normally are purchased ready mounted. The
244
Generation of
R
-F
best type mount is determined by the type crystal and its application, and usually an optimum mounting is furnished with the crystal.
However, certain features are desirable in all
holders. One of these is exclusion of moisture
and prevention of electrode oxidization. The
best means of accomplishing this is a metal
holder, hermetically sealed, with glass insulation and a metal -to -glass bond. However, such
holders are more expensive, and a ceramic or
phenolic holder with rubber gasket will serve
where requirements are not too exacting.
Temperature Control;
Crystal Ovens
Where the frequency tolerance requirements are
not too stringent and the
ambient temperature does not include extremes,
an AT -cut plate, or a BT -cut plate with optimum (mean temperature) turning point, will
often provide adequate stability without resorting to a temperature controlled oven. However, for broadcast stations and other applications where very close tolerances must be
maintained, a thermostatically controlled oven,
adjusted for a temperature slightly higher than
the highest ambient likely to be encountered,
must of necessity be employed.
vibrating string can
vibrate on its harmonics, a quartz crystal will
Harmonic Cut
Just as
Crystals
be made to
THE
Energy
RADIO
EXCITATION
6J5
ETC
EXCITATION
+5
loo -isov.
BASIC PIERCE" OSCILLATOR
HOT -CATHODE -PIERCE"
OSCILLATOR
Figure 6
THE PIERCE CRYSTAL OSCILLATOR
CIRCUIT
Shown at (A) is the basic Pierce crystal oscillator circuit. A capacitance of 10 to 75
µtd. normally will be required at C1 for
optimum operation. If a plate supply voltage
higher thon indicated is to be used, RFC'
may be replaced by a 22,000 -ohm 2-watt resistor. Shown at (B) is an alternative arrangement with the r -f ground moved to the
plate, and with the cathode floating. This
alternative circuit has the advantage that
the full r-f voltage developed across the
crystal may be used os excitation to the next
stage, since one side of the crystal is
grounded.
a
exhibit mechanical résonance (and therefore
electrical resonance) at harmonics of its funda-
mental frequency. When employed in the usual
holder, it is possible to excite the crystal
only on its odd harmonics (overtones).
By grinding the crystal especially for harmonic operation, it is possible to enhance its
operation as a harmonic resonator. BT and AT
cut crystals designed for optimum operation
on the 3d, 5th and even the 7th harmonic are
available. The 5th and 7th harmonic types,
especially the latter, require special holder
and oscillator circuit precautions for satisfactory operation, but the 3d harmonic type
needs little more consideration than a regular
fundamental type. A crystal ground for optimum
operation on a particular harmonic may or may
not be a good oscillator on a different harmonic or on the fundamental. One interesting
characteristic of a harmonic cut crystal is that
its harmonic frequency is not quite an exact
multiple of its fundamental, though the disparity is very small.
The harmonic frequency for which the crystal was designed is the working frequency. It
is not the fundamental since the crystal itself
actually oscillates on this working frequency
when it is functioning in the proper manner.
When a harmonic -cut crystal is employed, a
selective tuned circuit must be employed somewhere in the oscillator in order to discrimi-
nate against the fundamental frequency or undesired harmonics. Otherwise the crystal might
not always oscillate on the intended frequency.
For this reason the Pierce oscillator, later
described in this chapter, is not suitable for
use with harmonic -cut crystals, because the
only tuned element in this oscillator circuit
is the crystal itself.
For a given crystal operating as an anti -resonant tank in a given oscillator at fixed load impedance and plate and
screen voltages, the r -f current through the
crystal will increase as the shunt capacitance
C2 of figure 4 is increased, because this effectively increases the step -up ratio of C, to Cr.
the crystal
For a given shunt capacitance,
current for a given crystal is directly proportional to the r -f voltage across C,. This voltage may be measured by means of a vacuum
tube voltmeter having a low input capacitance,
and such a measurement is a more pertinent
one than a reading of r -f current by means of
a thermogalvanometer inserted in series with
one of the leads to the crystal holder.
The function of a crystal is to provide accurate frequency control, and unless it is used
in such a manner as to take advantage of its
inherent high stability, there is no point in
using a crystal oscillator. For this reason a
Crystal Current;
Heating and Fracture
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C
Crystal Oscillators
HANDBOOK
crystal oscillator should not be run at high
plate input in an attempt to obtain considerable power directly out of the oscillator, as
such operation will cause the crystal to heat,
with resultant frequency drift and possible
fracture.
13 -3
Crystal Oscillator
Circuits
Considerable confusion exists as to nomenclature of crystal oscillator circuits, due to a
tendency to name a circuit after its discoverer.
Nearly all the basic crystal oscillator circuits
were either first used or else developed independently by G. W. Pierce, but he has not been
so credited in all the literature.
Use of the crystal oscillator in master oscillator circuits in radio transmitters dates
back to about 1924 when the first application
articles appeared.
The Pierce The circuit of figure6A is the simplest crystal oscillator circuit. It
Oscillator
is one of those developed by
Pierce, and is generally known among amateurs as the Pierce oscillator. The crystal
simply replaces the tank circuit in a Colpitts
or ultra-audion oscillator. The r -f excitation
voltage available to the next stage is low, being somewhat less than that developed across
the crystal. Capacitor C, will make more of
the voltage across the crystal available for
excitation, and sometimes will be found necessary to ensure oscillation. Its value is small,
usually approximately equal to or slightly
greater than the stray capacitance from the
plate circuit to ground (including the grid of
the stage being driven).
If the r -f choke has adequate inductance, a
crystal (even a harmonic cut crystal) will almost invariably oscillate on its fundamental.
The Pierce oscillator therefore cannot be used
with harmonic cut crystals.
The circuit at (B) is the same as that of
(A) except that the plate instead of the cathode is operated at ground r -f potential. All of
the r -f voltage developed across the crystal
is available for excitation to the next stage,
but still is low for reasonable values of crystal current. For best operation a tube with low
heater - cathode capacitance is required. Excitation for the next stage may also be taken
from the cathode when using this circuit.
The circuit shown in fig ure 7A is also one used by
Pierce, but is more widely
referred to as the " \filler "oscillator. To avoid
Tuned -Plate
Crystal Oscillator
245
confusion, we shall refer to it as the tuned plate crystal oscillator. It is essentially an
Armstrong or tuned plate -tuned grid oscillator
with the crystal replacing the usual L -C grid
tank. The plate tank must be tuned to a frequency slightly higher than the anti -resonant
(parallel resonant) frequency of the crystal.
Whereas the Pierce circuits of figure 6 will
oscillate at (or very close to) the anti -resonant frequency of the crystal, the circuits of
figure 7 will oscillate at a frequency a little
above the anti -resonant frequency of the
crystal.
The diagram shown in figure 7A is the basic
circuit. The most popular version of the tuned plate oscillator employs a pentode or beam
tetrode with cathode bias to prevent excessive
plate dissipation when the circuit is not oscillating. The cathode resistor is optional. Its
omission will reduce both crystal current and
oscillator efficiency, resulting in somewhat
more output for a given crystal current. The
tube usually is an audio or video beam pentode
or tetrode, the plate -grid capacitance of such
tubes being sufficient to ensure stable oscillation but not so high as to offer excessive feedback with resulting high crystal current. The
6AG7 makes an excellent all- around tube for
this type circuit.
The usual type of crystal controlled h -f transmitter
Oscillator Circuits operates, at least part of
the time, on a frequency
which is an integral multiple of the operating
frequency of the controlling crystal. Hence,
oscillator circuits which are capable of providing output on the crystal frequency if desired, but which also can deliver output energy
on harmonics of the crystal frequency have
come into wide use. Four such circuits which
have found wide application are illustrated in
figures 7C, 7D, 7E, and 7F.
The circuit shown in figure 7C is recommended for use with harmonic -cut crystals
when output is desired on a multiple of the
oscillating frequency of the crystal. As an
example, a 25-Mc. harmonic-cut crystal may
be used in this circuit to obtain output on 50
Mc., or a 48 -Mc. harmonic-cut crystal may be
used to obtain output on the 144-Mc. amateur
band. The circuit is not recommended for use
with the normal type of fundamental- frequency
crystal since more output with fewer variable
elements can be obtained with the circuits of
7D and 7F.
The Pierce -harmonic circuit shown in figure 7D is satisfactory for many applications
which require very low crystal current, but
has the disadvantage that both sides of the
crystal are above ground potential. The Tri -tet
circuit of figure 7E is widely used and can
Pentode
Harmonic Crystal
246
Generation of
6J5,Erc
F
R
-F
6V6. 6AQ5,
RADIO
THE
Energy
6 AG7, 6AQ5.
ETC
3F
5763
002
150V.
+250
BASIC TUNED -PLATE OSCILLATOR
6AG7
F,
2F, 3F
+250 V.
V.
SPECIAL C RCUIT FOR USE WITH
HARMONIC CUT CRYSTAL.
RECOMMENDED TUNED -PLATE
OSCILLATOR
6AG7
F,
2F, 3F,
6AG7
4F
F.
2F, 3F,
4F
10111JF
150 ULF
22
+250
K
PIERCE HARMONIC CIRCUIT
"TRITET" CIRCUIT
Figure
V.
COLPITTS HARMONIC OSCILLATOR
7
COMMONLY USED CRYSTAL OSCILLATOR CIRCUITS
Shown at (A) is the basic tuned -plate crystal oscillator with a triode oscillator tube.
The plate tank must be tuned on the low -capacitance side of resonance to sustain
oscillation. (8) shows the tuned -plat oscillator as it is normally used with an a -f
power pentode to permit high output with relatively low crystal current.
Schematics (C), (D), (E), and (F) illustrate crystal oscillator circuits which can deliver moderate output energy on harmonics of the oscillating frequency of the crystal. (C) shows a special circuit which will permit use of a harmonic -cut crystal to
obtain output energy well into the v -h-f range. (D) is valuable when extremely low
crystal current is a requirement, but delivers relatively low output. (E) is commonly
used, but is subject to crystal damage if the cathode circuit is mistuned. (F) is
recommended as the most generally satisfactory from the standpoints of: low crystal current regardless of mis- adjustment, good output on harmonic frequencies, one
side of crystal is grounded, will oscillate with crystals from 1.5 to 70 Mc. without
adjustment, output tank may be tuned to the crystal frequency for fundamental output without stopping oscillation or changing frequency.
give excellent output with low crystal current.
However, the circuit has the disadvantages of
requiring a cathode coil, of requiring careful
setting of the variable cathode capacitor to
avoid damaging the crystal when changing frequency ranges, and of having both sides of
the crystal above ground potential.
The Colpitts harmonic oscillator of figure
7F is recommended as being the most generally satisfactory harmonic crystal oscillator
circuit since it has the following advantages:
(1) the circuit will oscillate with crystals
over a very wide frequency range with no
change other than plugging in or switching in
the desired crystal; (2) crystal current is ex-
tremely low; (3) one side of the crystal is
grounded, which facilitates crystal- switching
circuits; (4) the circuit will operate straight
through without frequency pulling, or it may
be operated with output on the second, third,
or fourth harmonic of the crystal frequency.
Crystal Oscillator
Tuning
The tunable circuits of all
oscillators
illustrated
should be tuned for maximum output as indicated by maximum excitation to the following stage, except that the
oscillator tank of tuned -plate oscillators (figure 7A and figure 7B) should be backed off
slightly towards the low capacitance side from
www.americanradiohistory.com
HANDBOOK
All -band Crystal Oscillator
6AG7
77. a 305
eaW M /N/JUC TOR
(2 0 vN)
+300
V.
NOTES
f"
I. Li'/sUN (2
OF eew 0301s)
(/" OF 80W .3003)
3. FOR 160 METER OPERATION ADD s 4/1/F. CONDENSER
BETWEEN PINS 418 OF 6A67. PLATE COIL= 35 2/P+.
2. L2 = /.gLN
(21-.0; Dew R awe)
4.
X' 7 MC.
CRYSTAL FOR HARMON /C OPERATION
Figure
8
ALL -3AND 6AG7 CRYSTAL OSCILLATOR
CAPABLE OF DRIVING
BEAM PENTODE TUBE
maximum output, as the oscillator then is in
a more stable condition and sure to start immediately when power is applied. This is especially important when the oscillator is keyed,
as for break-in c -w operation.
It is desirable to keep stray
shunt capacitances in the
crystal circuit as low as possible, regardless
of the oscillator circuit. If a selector switch
is used, this means that both switch and crystal sockets must be placed close to the oscillator tube socket. This is especially true of
harmonic -cut crystals operating on a comparatively high frequency. In fact, on the highest
frequency crystals it is preferable to use a
turret arrangement for switching, as the stray
capacitances can be kept lower.
Crystal Switching
Crystol Oscillator When the crystal
oscillator
is keyed, it is necessary
that crystal activity and oscillator -tube transconductance be moderately
high, and that oscillator loading and crystal
shunt capacitance be low. Below 2500 kc. and
above 6 Mc. these considerations become especially important. Keying of the plate voltage
(in the negative lead) of a crystal oscillator,
with the screen voltage regulated at about
150 volts, has been found to give satisfactory
results.
Keying
Versatile 6AG7
Crystal Oscillator
A
The 6AG7 tube may be
used in a modified Tri -tet
crystal oscillator, capable
of delivering sufficient power on all bands
247
from 160 meters through 10 meters to fully
drive a pentode tube, such as the 807, 2E26
or 6146. Such an oscillator is extremely useful for portable or mobile work, since it combines all essential exciter functions in one
tube. The circuit of this oscillator is shown
in figure 8. For 160, 80 and 40 meter operation the 6AG7 functions as a tuned-plate oscillator. Fundamental frequency crystals are
used on these three bands. For 20, 15 and 10
meter operation the 6AG7 functions as a Tritet oscillator with a fixed -tuned cathode circuit. The impedance of this cathode circuit
does not affect operation of the 6AG7 on the
lower frequency bands so it is left in the circuit at all times. A 7 -Mc. crystal is used for
fundamental output on 40 meters, and for harmonic output on 20, 15 and 10 meters. Crystal
current is extremely low regardless of the output frequency of the oscillator. The plate circuit of the 6AG7 is capable of tuning a frequency range of 2:1, requiring only two output
coils: one for 80-40 meter operation, and one
for 20, 15 and 10 meter operation. In some
cases it may be necessary to add 5 micromicrofarads of external feedback capacity between
the plate and control grid of the 6AG7 tube to
sustain oscillation with sluggish 160 meter
crystals.
Triode Overtone
Oscillators
The recent development of
reliable overtone crystals
capable of operation on the
third, fifth, seventh (or higher) overtones has
made possible v -h -f output from a low frequency crystal by the use of a double triode regenerative oscillator circuit. Some of the new
twin triode tubes such as the 12AU7, 12AV7
and 6J6 are especially satisfactory when used
in this type of circuit. Crystals that are ground
for overtone service may be made to oscillate
on odd overtone frequencies other than the
one marked on the crystal holder. A 24 -Mc.
overtone crystal, for example, is a specially
ground 8 -Mc. crystal operating on its third
overtone. In the proper circuit it may be made
to oscillate on 40 Mc. (fifth overtone), 56 Mc.
(seventh overtone), or 72 Mc. (ninth overtone).
Even the ordinary 8 -Mc. crystals not designed
for overtone operation may be made to oscillate readily on 24 Mc. (third overtone) in these
circuits.
A variety of overtone oscillator circuits is
shown in figure 9. The oscillator of figure 9A
is attributed to Frank Jones, W6AJF. The first
section of the 6J6 dual triode comprises a regenerative oscillator, with output on either the
third or fifth overtone of the crystal frequency.
The regenerative loop of this oscillator consists of a condenser bridge made up of C, and
C2, with the ratio C2 /C, determining the amount
of regenerative feedback in the circuit. With
www.americanradiohistory.com
248
Generation of
R
RADIO
THE
Energy
-F
2F 6J6
4F
6,9,10 o915 F
FOR 7 MC. CRYSTAL
+300
+300
AO
LI =28T e 24 ON NATIONAL
V
FORM
L 2. $ T Il
FORM
V.
©
JONES HARMONIC OSCILLATOR
12AÚ7
XPSO
ON NATIONAL XRSO
COLPITTS HARMONIC OSCILLATOR
6J6
3F
6, 9F
3F
/!
/s0
50
6,9F
RFC
FOR
+300
©
v.
L I= 9 7.
L2. 4
T
MC. CRYSTAL
a
5003 9iW M /N /DUCTOR
13003 817 MIN/DOCTOR
L
+300V
I
=
FOR 8MC. CRYSTAL
M NIDUC TOR,
TAP AT 37. FROM GRID ENO
/07 a 30 /I 64W
THESE COLS MADE FROM SINGLE SECT /ON
OF
NE TRN BROKEN TO
O /V /OE IINDUCTOR INTO TWO COILS S
REGENERATIVE HARMONIC OSCILLATOR
OD
REGENERATIVE HARMONIC OSCILLATOR
- 12AT7 (or 6AB4)
12AT7
6,9F
_
_
!00
F=IA4MC.
1_
L2
+200
LI=STe/E,
L2=/ T.
+300
E©
V.
f-
O.SPACED
NOOXUPWIRE,
O.
V.
CATHODE FOLLOWER OVERTONE OSCILLATOR
VARIOUS TYPES
OF
OVERTONE
OF
V.H.F. OVERTONE OSCILLATOR
Figure 9
OSCILLATORS USING
TUBES
an 8 -plc. crystal, output from the first section
of the 6J6 tube may be obtained on either 24
Mc. or 40 Mc., depending upon the resonant
frequency of the plate circuit inductor, L,. The
second half of the 636 acts as a frequency
multiplier, its plate circuit, L2, tuned to the
sixth or ninth harmonic frequency when L, is
tuned to the third overtone, or to the tenth
harmonic frequency when L, is tuned to the
fifth overtone.
Figure 9B illustrates a Colpitts overtone
oscillator employing a 636 tube. This is an
outgrowth of the Colpitts harmonic oscillator
of figure 7F. The regenerative loop in this
MINIATURE
case consists of
C
DOUBLE -TRIODE
C2 and RFC between the
grid, cathode and ground of the first section
of the 6J6. The plate circuit of the first section is tuned to the second overtone of the
crystal, and the second section of the 636
doubles to the fourth harmonic of the crystal.
This circuit is useful in obtaining 28 -Mc. output from a 7 -Mc. crystal and is highly popular
in mobile work.
The circuit of figure 9C shows a typical regenerative overtone oscillator employing a
12AÚ7 double triode tube. Feedback is controlled by the number of turns in L2, and the
coupling between L2 and L,. Only enough feed-
www.americanradiohistory.com
HANDBOOK
R
back should be employed to maintain proper
oscillation of the crystal. Excessive feedback
will cause the first section of the 12ÁU7 to
oscillate as
a
self- excited TNT oscillator,
independent of the crystal. A variety of this
circuit is shown in figure 9D, wherein a tapped
coil,
is used in place of the two separate
coils. Operation of the circuit is the same in
either case, regeneration now being controlled
by the placement of the tap on L,.
A cathode follower overtone oscillator is
shown in figure 9E. The cathode coil,
is
chosen so as to resonate with the crystal and
tube capacities just below the third overtone
frequency of the crystal. For example, with an
8 -Mc. crystal, L3 is tuned to 24 Mc.. L, resonates with the circuit capacities to 23.5 Mc.,
and the harmonic tank circuit of the second
section of the 12AT7 is tuned either to 48 Mc.
or 72 11c. If a 24 -Mc. overtone crystal is used
in this circuit, L, may be tuned to 72 Mc., L,
resonates with the circuit capacities to 70
Mc., and the harmonic tank circuit,
is tuned
to 144 Mc. If there is any tendency towards
self-oscillation in the circuit, it may be eliminated by a small amount of inductive coupling
between L2 and L3. Placing these coils near
each other, with the winding of L, correctly
polarized with respect to L3 will prevent selfoscillation of the circuit.
The use of a 144 -Mc. overtone crystal is
illustrated in figure 9F. A 6AB4 or one -half
of a 12AT7 tube may be used, with output
directly in the 2 -meter amateur band. A slight
amount of regeneration is provided by the one
which is loosely coupled to the
turn link,
144 -Mc. tuned tank circuit, L, in the plate circuit of the oscillator tube. If a 12AT7 tube
and a 110 -Mc. crystal are employed, direct output in the 220 -Mc. amateur band may be obtained from the second half of the 12AT7.
L
L
L
L
13 -4
Radio Frequency
Amplifiers
The output of the oscillator stage in a transmitter (whether it be self-controlled or crystal
controlled) must be kept down to a fairly low
level to maintain stability and to maintain a
factor of safety from fracture of the crystal
when one is used. The low power output of
the oscillator is brought up to the desired
power level by means of radio -frequency amplifiers. The two classes of r -f amplifiers that
find widest application in radio transmitters
are the Class B and Class C types.
The Class B
Amplifier
Class B amplifiers are used in a
radio -telegraph transmitter when
maximum power gain and mini-
-F
Amplifiers
249
output is desired in a particular stage. A Class B amplifier operates with
cutoff bias and a comparatively small amount
of excitation. Power gains of 20 to 200 or so
are obtainable in a well- designed Class B
amplifier. The plate efficiency of a Class B
c -w amplifier will run around 65 per cent.
mum harmonic
Another type of Class B ampli fier is the Class B linear stage
as employed in radiophone work.
This type of amplifier is used to increase the
level of a modulated carrier wave, and depends for its operation upon the linear relation between excitation voltage and output
voltage. Or, to state the fact in another manner, the power output of a Class B linear stage
varies linearly with the square of the excitation voltage.
The Class B linear amplifier is operated
with cutoff bias and a small value of excitation, the actual value of exciting power being
such that the power output under carrier conditions is one -fourth of the peak power capabilities of the stage. Class B linears are very
widely employed in broadcast and commercial
installations, but are comparatively uncommon
in amateur application, since tubes with high
plate dissipation are required for moderate
output. The carrier efficiency of such an amplifier will vary from approximately 30 per
cent to 35 per cent.
The Class B
Linear
Class C amplifiers are very wide ly used in all types of transmitters. Good power gain may be
obtained (values of gain from 3 to 20 are common) and the plate circuit efficiency may be,
under certain conditions, as high as 85 per
cent. Class C amplifiers operate with considerably more than cutoff bias and ordinarily with
a large amount of excitation as compared to a
Class B amplifier. The bias for a normal Class
C amplifier is such that plate current on the
stage flows for approximately 120° of the 360°
excitation cycle. Class C amplifiers are used
in transmitters where a fairly large amount of
excitation power is available and good plate
circuit efficiency is desired.
The Class C
Amplifier
The characteristic of a Class
C amplifier which makes it
linear with respect to
changes in plate voltage is that which allows
such an amplifier to be plate modulated for
radiotelephony. Through the use of higher bias
than is required for a c -w Class C amplifier
and greater excitation, the linearity of such
an amplifier may be extended from zero plate
voltage to twice the normal value. The output
power of a Class C amplifier, adjusted for
plate modulation, varies with the square of the
Plate Modulated
Class C
www.americanradiohistory.com
Generation
250
of
R
-F
THE
Energy
plate voltage. This is the same condition that
would take place if a resistor equal to the
voltage on the amplifier, divided by its plate
current, were substituted for the amplifier.
Therefore, the stage presents a resistive load
RADIO
Excessive grid current damages tubes by
overheating the grid structure; beyond a certain point of grid drive, no increase in power
output can be obtained for a given plate voltage.
to the modulator.
Grid Modulated
If the grid current to a
Class
amplifier is reduced to a
low value, and the plate loading is increased to the point where the plate
dissipation approaches the rated value, such
an amplifier may be grid modulated for radiotelephony. If the plate voltage is raised to
quite a high value and the stage is adjusted
carefully, efficiencies as high as 40 to 43 per
cent with good modulation capability and comparatively low distortion may be obtained.
Fixed bias is required. This type of operation
is termed Class C grid-bias modulation.
Class
Grid Excitation
Adequate grid
excitation
must be available for Class
service. The excitation for a
plate- modulated Class C stage must be sufficient to produce a normal value of d -c grid current with rated bias voltage. The bias voltage
preferably should be obtained from a combination of grid leak and fixed C -bias supply.
Cutoff bias can be calculated by dividing
the amplification factor of the tube into the
d-c plate voltage. This is the value normally
used for Class B amplifiers (fixed bias, no
grid resistor). Class C amplifiers use from 1
to 5 times this value, depending upon the available grid drive, or excitation, and the desired
plate efficiency. Less grid excitation is needed for c -w operation, and the values of fixed
bias (if greater than cutoff) may be reduced, or
the value of the grid leak resistor can be lowered until normal rated d-c grid current flows.
The values of grid excitation listed for each
type of tube may be reduced by as much as
50 per cent if only moderate power output and
plate efficiency are desired. When consulting
the tube tables, it is well to remember that
the power lost in the tuned circuits must be
taken into consideration when calculating the
available grid drive. At very high frequencies,
the r -f circuit losses may even exceed the
power required for actual grid excitation.
Link coupling between stages, particularly
to the final amplifier grid circuit, normally will
provide more grid drive than can be obtained
from other coupling systems. The number of
turns in the coupling link, and the location of
the turns on the coil, can be varied with respect to the tuned circuits to obtain the greatest grid drive for allowable values of buffer
or doubler plate current. Slight readjustments
sometimes can be made after plate voltage
has been applied to the driver tube.
B or
Class
Neutralization of
R.F. Amplifiers
13 -5
C
C
C
The plate -to -grid feedback capacitance of
triodes makes it necessary that they be neutralized for operation as r -f amplifiers at frequencies above about 500 kc. Those screen grid tubes, pentodes, and beam tetrodes which
have a plate -to -grid capacitance of 0.1 µµEd.
or less may be operated as an amplifier without neutralization in a well -designed amplifier
up to 30 Mc.
Neutralizing
Circuits
The object of neutralization is
to cancel or neutralize the capacitive feedback of energy from
plate to grid. There are two general methods
by which this energy feedback may be eliminated: the first, and the most common method,
is through the use of a capacitance bridge,
and the second method is through the use of a
parallel reactance of equal and opposite polarity to the grid -to-plate capacitance, to nullify the effect of this capacitance.
Examples of the first method are shown in
figure 10. Figure l0A shows a capacity neutralized stage employing a balanced tank circuit. Phase reversal in the tank circuit is obtained by grounding the center of the tank coil
to radio frequency energy by condenser C.
Points A and B are 180 degrees out of phase
with each other, and the correct amount of out
of phase energy is coupled through the neutralizing condenser NC to the grid circuit of
the tube. The equivalent bridge circuit of this
is shown in figure 11A. It is seen that the
bridge is not in balance, since the plate -filament capacity of the tube forms one leg of the
bridge, and there is no corresponding capacity
from the neutralizing condenser (point B) to
ground to obtain a complete balance. In addition, it is mechanically difficult to obtain a
perfect electrical balance in the tank coil, and
the potential between point A and ground and
point B and ground in most cases is unequal.
This circuit, therefore, holds neutralization
over a very small operating range and unless
tubes of low interelectrode capacity are used
the inherent unbalance of the circuit will permit only approximate neutralization.
Split-Stator
Plate Neutralization
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Figure 10B shows the neu-
tralization circuit which is
most widely used in single ended r -f stages. The use of
HANDBOOK
Neutralization
Figure
COMMON
10
NEUTRALIZING CIRCUITS FOR SINGLE -ENDED AMPLIFIERS
split -stator plate capacitor makes the electrical balance of the circuit substantially independent of the mutual coupling within the coil
and also makes the balance independent of the
place where the coil is tapped. With conventional tubes this circuit will allow one neutralization adjustment to be made on, say, 28 Mc.,
and this adjustment usually will hold sufficiently close for operation on all lower frequency bands.
Condenser C, is used to balance out the
plate -filament capacity of the tube to allow a
perfect neutralizing balance at all frequencies.
The equivalent bridge circuit is shown in figure 11B. If the plate- filament capacity of the
tube is extremely low (100TH triode, for example), condenser C, may be omitted, or may
merely consist of the residual capacity of NC
to ground.
a
split grid tank circuit
also be used for neutralization of a triode tube as shown in figure
IOC. Out of phase voltage is developed across
a balanced grid circuit, and coupled through
NC to the single -ended plate circuit of the
tube. The equivalent bridge circuit is shown
in figure 11C. This circuit is in balance until
the stage is in operation when the loading effect of the tube upon one -half of the grid circuit throws the bridge circuit out of balance.
The amount of unbalance depends upon the
grid -plate capacity of the tube, and the amount
of mutual inductance between the two halves
Grid Neutralization
251
A
may
of the grid coil. If an r -f voltmeter is placed
between point A and ground, and a second
voltmeter placed between point B and ground
the loading effect of the tube will be noticeable. When the tube is supplied excitation
with no plate voltage, NC may be adjusted
until the circuit is in balance. When plate
voltage is applied to the stage, the voltage
from point A to ground will decrease, and the
voltage from point B to ground will increase,
both in direct proportion to the amount of circuit unbalance. The use of this circuit is not
recommended above 7 Mc., and it should be
used below that frequency only with low in-
ternal capacity tubes.
Two tubes of the same type
can be connected for push -pull
operation so as to obtain twice
as much output as that of a single tube. A
push -pull amplifier, such as that shown in figure 12 also has an advantage in that the circuit can more easily be balanced than a single tube r -f amplifier. The various inter -electrode
capacitances and the neutralizing capacitors
are connected in such a manner that the reactances on one side of the tuned circuits are
exactly equal to those on the opposite side.
For this reason, push -pull r -f amplifiers can
be more easily neutralized in very- high -frequency transmitters; also, they usually remain
in perfect neutralization when tuning the amplifier to different bands.
The circuit shown in figure 12 is perhaps
Push -Pull
Neutralization
Generation of
252
-F
R
THE
Energy
RADIO
F.AC
OA
BRIDGE EQUIVALENT OF FIGURE
IO -A
C
Figure 12
STANDARD CROSS- NEUTRALIZED
PUSH -PULL TRIODE AMPLIFIER
OB
BRIDGE EQUIVALENT OF FIGURE
10
-B
C
actance, coupling energy back from the plate
to the grid circuit. If this capacitance is paralleled with an inductance having the same
value of reactance of opposite sign, the reactance of one will cancel the reactance of
the other and a high -impedance tuned circuit
will result.
This neutralization circuit can be used on
ultra -high frequencies where other neutralization circuits are unsatisfactory. This is true
because the lead length in the neutralization
circuit is practically negligible. The circuit
can also be used with push -pull r -f amplifiers.
In this case, each tube will have its own neutralizing inductor connected from grid to plate.
The main advantage of this arrangement is
that it allows the use of single -ended tank
circuits with a single -ended amplifier.
The chief disadvantage of the shunt neutralized arrangement is that the stage must be reneutralized each time the stage is retuned to
a new frequency sufficiently removed that the
grid and plate tank circuits must be retuned to
resonance. However, by the use of plug -in
coils it is possible to change to a different
band of operation by changing the neutralizing coil at the same time that the grid and
plate coils are changed.
The 0.0001-pfd. capacitor in series with
the neutralizing coil is merely a blocking capacitor to isolate the plate voltage from the
grid circuit. The coil L will have to have a
very large number of turns for the band of operation in order to be resonant with the comparatively small grid -to -plate capacitance. But
since, in all ordinary cases with tubes operating on frequencies for which they were designed, the L/C ratio of the tuned circuit will
be very high, the coil can use comparatively
small wire, although it must be wound on air
or very low loss dielectric and must be insulated for the sum of the plate r-f voltage and
the grid r-f voltage.
from grid to plate
(RES.DUAL
CAPACITY)`
©
CG-r
(s1IALL)
RFC
BRIDGE EQUIVALENT OF FIGURE 10-G
Figure 11
EQUIVALENT NEUTRALIZING CIRCUITS
most commonly used arrangement for a
push -pull r -f amplifier stage. The rotor of the
grid capacitor is grounded, and the rotor of the
plate tank capacitor is by- passed to ground.
the
Shunt or Coil
Neutralization
The feedback of energy from
grid to plate in an unneutral-
ized r -f amplifier is
a
result of
the grid -to -plate capacitance of the amplifier
tube. A neutralization circuit is merely an
electrical arrangement for nullifying the effect
of this capacitance. All the previous neutralization circuits have made use of a bridge circuit for balancing out the grid -to -plate energy
feedback by feeding hack an equal amount of
energy of opposite phase.
Another method of eliminating the feedback
effect of this capacitance, and hence of neutralizing the amplifier stage, is shown in figure 13. The grid -to -plate capacitance in the
triode amplifier tube acts as a capacitive re-
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Neutralizing
HANDBOOK
Procedure
253
grid excitation is applied, even though no primary a-c voltage is being fed to the plate transformer.
further check on the neutralization of any
amplifier can be made by noting whether
maximum grid current on the stage comes at
the same point of tuning on the plate tuning
capacitor as minimum plate current. This check
is made with plate voltage on the amplifier
and with normal antenna coupling. As the plate
tuning capacitor is detuned slightly from resonance on either side the grid current on the
A
r -f
Figure
13
COIL NEUTRALIZED AMPLIFIER
This neutralization circuit is very effective
with triode tubes on any frequency, but is
particularly effective in the v -h-f range. The
coil L is adjusted so that it resonates at the
operating frequency with the grid -to -plate
capacitance of the tube. Capacitor C may be
a very small unit of the low- capacitance
neutralizing type and is used to trim the circuit to resonance at the operating frequency.
If some means of varying the inductance of
the coil a small amount is available, the
trimmer capacitor is not needed.
stage should decrease the same amount and
without any sudden jumps on either side of
resonance. This will be found to be a very
precise indication of accurate neutralization
in either a triode or beam -tetrode r -f amplifier
stage, so long as the stage is feeding a load
which presents a resistive impedance at the
operating frequency.
Push-pull circuits usually can be more completely neutralized than single -ended circuits
at very high frequencies. In the intermediate
range of from 3 to 15 Mc., single -ended circuits will give satisfactory results.
Neutralization of
Screen -Grid
13 -6
R
-F
Amplifiers
Neutralizing
Procedure
An r-f amplifier is neutralized to prevent
self-oscillation or regeneration. A neon bulb,
a flashlight lamp and loop of wire, or an r -f
galvanometer can be used as a null indicator
for neutralizing low -power stages. The plate
voltage lead is disconnected from the r-f amplifier stage while it is being neutralized.
Normal grid drive then is applied to the r -f
stage, the neutralizing indicator is coupled
to the plate coil, and the plate tuning capacitor is tuned to resonance. The neutralizing
capacitor (or capacitors) then can be adjusted
until minimum r.f. is indicated for resonant
settings of both grid and plate tuning capacitors. Both neutralizing capacitors are adjusted simultaneously and to approximately the
same value of capacitance when a physically
symmetrical push -pull stage is being neutralized.
A final check for neutralization should be
made with a d-c milliammeter connected in the
grid leak or grid -bias circuit. There will be
no movement of the meter reading as the plate
circuit is tuned through resonance (without
plate voltage being applied) when the stage
is completely neutralized.
Plate voltage should be completely removed
by actually opening the d-c plate circuit. If
there is a d-c return through the plate supply,
a small amount of plate current will flow when
Radio-frequency amplifiers
using screen -grid tubes can
be operated without any ad-
ditional provision for neutralization at frequencies up to about 15 Mc.,
provided adequate shielding has been provided
between the input and output circuits. Special
v -h -f screen -grid and beam tetrode tubes such
as the 2E26, 807W, and 5516 in the low -power
category and HK -257B, 4E27/8001, 4 -125A,
and 4 -250A in the medium -power category can
frequently be operated at frequencies as high
as 100 Mc. without any additional provision
for neutralization. Tubes such as the 807,
2E22, HY -69, and 813 can be operated with
good circuit design at frequencies up to 30
Mc. without any additional provision for neutralization. The 815 tube has been found to
require neutralization in many cases above
20 Mc., although the 829B tube will operate
quite stably at 100 Mc. without neutralization.
None of these tubes, however, has perfect
shielding between the grid and the plate, a
condition brought about by the inherent inductance of the screen leads within the tube
itself. In addition, unless "watertight" shielding is used between the grid and plate circuits
of the tube a certain amount of external leakage between the two circuits is present. These
difficulties may not be serious enough to require neutralization of the stage to prevent
oscillation, but in many instances they show
up in terms of key -clicks when the stage in
question is keyed, or as parasitics when the
stage is modulated. Unless the designer of the
equipment can carefully check the tetrode
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254
Generation of
R
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Energy
Figure
RADIO
14
NEUTRALIZING CIRCUITS FOR
BEAM TETRODES
conventional cross neutralized circuit for use with push-pull beam tetrodes is
shown at (A). The neutralizing capacitors (NC) usually consist of small plates
or
rods mounted alongside the plate elements of the tubes. (B) and (C) show "grid
neutralized" circuits for use with a single -ended tetrode stage having either link
coupling or capacitive coupling into the grid tank. (D) shows a method of tuning the
screen -lead inductance to accomplish neutralization in a single -frequency -h-f
tetrode amplifier, while (E) shows a method of neutralization by increasing the vgrid
to -plate capacitance on a tetrode when the operating frequency is higher than
that
frequency where the tetrode is "self- neutralized" as a result of series resonance
in the screen lead. Methods (D) and (E) normally are not practicable at frequencies
below about 50 Mc. with the usual types of beam tetrode tubes.
A
stage for miscellaneous feedback between the
grid and plate circuits, and make the necessary circuit revisions to reduce this feedback
to an absolute minimum, it is wise to neutralize the tetrode just as if it were a triode tube.
In most push -pull tetrode amplifiers the simplest method of accomplishing neutralization
is to use the cross -neutralized capacitance
bridge arrangement as normally employed with
triode tubes. The neutralizing capacitances,
however, must be very much smaller than used
with triode tubes, values of the order of 0.2
wifd. normally being required with beam tetrode
tubes. This order of capacitance is far less
than can be obtained with a conventional neutralizing capacitor at minimum setting, so the
neutralizing arrangement is most commonly
made especially for the case at hand. Most
common procedure is to bring a conductor (connected to the opposite grid) in the vicinity of
the plate itself or of the plate tuning capacitor
of one of the tubes. Either one or two such
capacitors may be used, two being normally
used on a higher frequency amplifier in order
to maintain balance within the stage.
An example of this is shown in figure 14A.
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H
Tetrode Neutralization
A N D B O O K
Neutralizing
single -ended tetrode r -f amplifier stage may be neutral ized in the same manner as
illustrated for a push -pull
stage in figure 14A, provided a split- stator
tank capacitor is in use in the plate circuit.
However, in the majority of single -ended tetrode r -f amplifier stages a single- section capacitor is used in the plate tank. Hence, other
neutralization procedures must be employed
when neutralization is found necessary.
The circuit shown in figure 14B is not a
true neutralizing circuit, in that the plate -togrid capacitance is not balanced out. However,
the circuit can afford the equivalent effect by
isolating the high resonant impedance of the
grid tank circuit from the energy fed back from
plate to grid. When NC and C are adjusted to
bear the following ratio to the grid -to-plate
capacitance and the total capacitance from
A
Single -Ended
Tetrode Stages
grid -to- ground in the output tube:
NC
CsP
C
Cas
both ends of the grid tank circuit will be at the
same voltage with respect to ground as a result
of r -f energy fed back to the grid circuit. This
means that the impedance from grid to ground
will be effectively equal to the reactance of
the grid -to- cathode capacitance in parallel
with the stray grid -to- ground capacitance, since
the high resonant impedance of the tuned circuit in the grid has been effectively isolated
from the feedback path. It is important to note
that the effective grid -to- ground capacitance
of the tube being neutralized includes the
rated grid -to- cathode or input capacitance of
the tube, the capacitance of the socket, wiring
capacitances and other strays, but it does not
include the capacitances associated with the
grid tuning capacitor. Also, if the tube is being excited by capacitive coupling from a preceding stage (as in figure 14C), the effective
grid-to- ground capacitance includes the output capacitance of the preceding stage and
its associated socket and wiring capacitances.
The provisions discussed in
the previous paragraphs are
for neutralization of the small,
though still important at the
higher frequencies, grid -to -plate capacitance
of beam -tetrode tubes. However, in the vicinity
of the upper- frequency limit of each tube type
the inductance of the screen lead of the tube
becomes of considerable importance. With a
tube operating at a frequency where the inductance of the screen lead is appreciable,
the screen will allow a considerable amount
of energy leak- through from plate to grid even
Cancellation of
Screen -Lead
Inductance
255
though the socket terminal on the tube is carefully by- passed to ground. This condition takes
place even though the socket pin is bypassed
since the reactance of the screen lead
will allow a moderate amount of r-f potential
to appear on the screen itself inside the electrode assembly in the tube. This effect has
been reduced to a very low amount in such
tubes as the Hytron 5516, and the Eimac 4X150A
and 4X500A but it is still quite appreciable in
most beam -tetrode tubes.
The effect of screen -lead inductance on the
stability of a stage can be eliminated at any
particular frequency by one of two methods.
These methods are: (1) Tuning out the screen lead inductance by series resonating the screen
lead inductance with a capacitor to ground.
This method is illustrated in figure 14D and is
commonly employed in commercially -built equipment for operation on a narrow frequency band
in the range above about 75 Mc. The other
method (2) is illustrated in figure 14E and
consists in feeding back additional energy
from plate to grid by means of a small capacitor connected between these two elements.
Note that this capacitor is connected in such
a manner as to increase the effective grid -toplate capacitance of the tube. This method
has been found to be effective with 807 tubes
in the range above 50 Mc. and with tubes such
as the 4 -125A and 4 -250A in the vicinity of
their upper frequency limits.
Note that both these methods of stabilizing
a beam-tetrode v -h -f amplifier stage by cancellation of screen -lead inductance are suitable only for operation over a relatively narrow
band of frequencies in the v -h-f range. At lower frequencies both these expedients for reducing the effects of screen -lead inductance
will tend to increase the tendency toward oscillation of the amplifier stage.
stage cannot be completely neutralized, the difficulty
usually can be traced to one or
more of the following causes: (1) Filament
leads not by- passed to the common ground of
that particular stage. (2) Ground lead from the
rotor connection of the split -stator tuning capacitor to filament open or too long. (3) Neutralizing capacitors in a field of excessive
r.f. from one of the tuning coils. (4) Electromagnetic coupling between grid and plate
coils, or between plate and preceding buffer
or oscillator circuits. (5) Insufficient shielding
or spacing between stages, or between grid
and plate circuits in compact transmitters.
(6) Shielding placed too close to plate circuit
coils, causing induced currents in the shields.
(7) Parasitic oscillations when plate voltage
is applied. The cure for the latter is mainly a
matter of cut and try -rearrange the parts,
Neutralizing
Problems
When a
-
256
Generation of
R
-F
THE
Energy
GRID
LEAK
RADIO
OUT
INTERWOUND COILS
(UNITY COUPLING)
Figure
CONVENTIONAL
16
TRIODE
FREQUENCY
MULTIPLIER
Figure
15
GROUNDED -GRID AMPLIFIER
This type of triode amplifier requires no
neutralization, but can be used only with
tubes having o relatively low plate -to- cathode
capacitance
change the length of grid or plate or neutralizing leads, insert a parasitic choke in the grid
lead or leads, or eliminate the grid r -f chokes
which may be the cause of a low- frequency
Small triodes such as the 604 operate satisfactorily as frequency multipliers, and can
deliver output well into the v -h -t ronge. Resistor R normally will have a value in the
vicinity of 100,000 ohms.
given output, because a moderate amount of
power is delivered to the amplifier load by the
driver stage of a grounded -grid amplifier.
13 -8
Frequency Multipliers
parasitic(in conjunction with plate r -f chokes).
13- 7
Grounded Grid
Amplifiers
Certain triodes have
grid configuration
results in very low
plate to filament capacitance when the control
grid is grounded, the grid acting as an effective shield much in the manner of the screen
in a screen -grid tube.
By connecting such a triode in the circuit of
figure 15, taking the usual precautions against
stray capacitive and inductive coupling between input and output leads and components,
a stable power amplifier is realized which requires no neutralization.
At ultra -high frequencies, where it is difficult to obtain satisfactory neutralization with
conventional triode circuits (particularly when
a wide band of frequencies is to be covered),
the grounded -grid arrangement is about the only
practicable means of employing a triode ama
and lead arrangement which
plifier.
Because of the large amount of degeneration
inherent in the circuit, considerably more excitation is required than if the same tube were
employed in a conventional grounded- cathode
circuit. The additional power required to drive
a triode in a grounded -grid amplifier is not
lost, however, as it shows up in the output circuit and adds to the power delivered to the
load. But nevertheless it means that a larger
driver stage is required for an amplifier of
Quartz crystals and variable- frequency oscillators are not ordinarily used for direct control of the output of high- frequency transmitters. Frequency multipliers are usually employed to multiply the frequency to the desired
value. These multipliers operate on exact multiples of the excitation frequency; a 3.6-Mc.
crystal oscillator can be made to control the
output of a transmitter on 7.2 or 14.4 Mc., or
on 28.8 Mc., by means of one or more frequency
multipliers. Chen used at twice frequency,
they are often termed frequency doublers. A
simple doubler circuit is shown in figure 16.
It consists of a vacuum tube with its plate circuit tuned to twice the frequency of the grid
driving circuit. This doubler can be excited
from a crystal oscillator or another multiplier
or amplifier stage.
Doubling is best accomplished by operating
the tube with high grid bias. The grid circuit
is driven approximately to the normal value of
d -c grid current through the r-f choke and grid leak resistor, shown in figure 16. The resistance value generally is from two to five times
as high as that used with the same tube for
straight amplification. Consequently, the grid
bias is several times as high for the same
value of grid current.
Neutralization is seldom necessary in a
doubler circuit, since the plate is tuned to
twice the frequency of the grid circuit. The
impedance of the grid driving circuit is very
low at the doubling frequency, and thus there
is little tendency for self -excited oscillation.
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HANDBOOK
Frequency Multipliers
257
TANN CIRCUIT OUTPUT VOLTAGE
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Figure 18
ILLUSTRATING THE ACTION OF
A
FREQUENCY DOUBLER
degrees or less. Under these conditions
the efficiency will be on the same order as
the reciprocal of the harmonic on which the
stage operates. In other words the efficiency
of a doubler will be approximately % or 50
per cent, the efficiency of a tripler will be
approximately / or 33 per cent and that of a
quadrupler will be about 25 per cent. With good
stage design the efficiency can be somewhat
greater than these values, but as the angle
of flow is made greater than these limiting
values, the efficiency falls off rapidly. The
reason is apparent from a study of figure 18.
The pulses ABC, EFG, JKL illustrate 180 degree excitation pulses under Class B operation, the solid straight line indicating cutoff
bias. If the bias is increased by N times, to
the value indicated by the dotted straight line,
and the excitation increased until the peak
r -f voltage with respect to ground is the same
as before, then the excitation frequency can
be cut in half and the effective excitation
pulses will have almost the same shape as
before. The only difference is that every other
pulse is missing; MNO simply shows where
the missing pulse would go. However, if the
Q of the plate tank circuit is high, it will have
sufficient flywheel effect to carry over through
the missing pulse, and the only effect will be
that the plate input and r -f output at optimum
loading drop to approximately half. As the input frequency is half the output frequency, an
efficient frequency doubler is the result.
By the same token, a tripler or quadrupler
can be analyzed, the tripler skipping two excitation pulses and the quadrupler three. In
each case the excitation pulse ideally should
be short enough that it does not exceed 180
degrees at the output frequency; otherwise the
excitation actually is bucking the output over
a portion of the cycle.
In actual practice, it is found uneconomical
to provide sufficient excitation to run a tripler
or quadrupler in this fashion. Usually the ex45
Figure 17
FREQUENCY MULTIPLIER CIRCUITS
The output of a triode v-h -f frequency multiplier often maybe increased by neutralization
of the grid-to -plate capacitance as shown at
(A) above. Such o stage also may be operated as a straight amplifier when the occasion demands. A pentode frequency multiplier is shown at (B). Conventional power
tetrodes operate satisfactorily as multipliers
so long as the output frequency is below
about 100 Mc. Above this frequency special
v -h -f tetrodes must be used to obtain satisfactory output.
Frequency doublers require bias of several
times cutoff; high -ft tubes therefore are desirable for this type of service. Tubes which
have amplification factors from 20 to 200 are
suitable for doubler circuits. Tetrodes and
pentodes make excellent doublers. Low -ft
triodes, having amplification constants of from
3 to 10, are not applicable for doubler service.
In extreme cases the grid voltage must be as
high as the plate voltage for efficient doubling
action.
Angle of Flow
in Frequency
The angle of plate current flow
in a frequency multiplier is a
Multipliers
very important factor in determining the efficiency. As the
angle of flow is decreased for a given value
of grid current, the efficiency increases. To
reduce the angle of flow, higher grid bias is
required so that the grid excitation voltage
will exceed the cutoff value for a shorter portion of the exciting -voltage cycle. For a high
order of efficiency, frequency doublers should
have an angle of flow of 90 degrees or less,
tripiers 60 degrees or less, and quadruplers
www.americanradiohistory.com
Generation of
258
Figure
R
-F
THE
Energy
RADIO
19
Figure
PUSH -PUSH
FREQUENCY DOUBLER
The output of o doubler stage may be materially increased through the use of a push -push
circuit such as illustrated above.
citation pulses will be at least 90 degrees at
the exciting frequency, with correspondingly
low efficiency, but it is more practicable to
accept the low efficiency and build up the output in succeeding amplifier stages. The efficiency can become quite low before the power
gain becomes less than unity.
Two tubes can be connected in
parallel to give twice the output
of a single -tube doubler. If the
grids are driven out of phase instead of in
phase, the tubes then no longer work simultaneously, but rather one at a time. The effect
is to fill in the missing pulses (figure 18).
Not only is the output doubled, but several
advantages accrue which cannot be obtained
by straight parallel operation.
Chief among these is the effective neutralization of the fundamental and all odd harmonics, an advantage when spurious emissions
must be minimized. Another advantage is that
when the available excitation is low and excitation pulses exceed 90 degrees, the output
and efficiency will be greater than for the
same tubes connected in parallel.
The same arrangement may be used as a
quadrupler, with considerably better efficiency
than for straight parallel operation, because
seldom is it practicable to supply sufficient
excitation to permit 45 degree excitation
pulses. As pointed out above, the push -push
arrangement exhibits better efficiency than a
single ended multiplier when excitation is inadequate for ideal multiplier operation.
A typical push -push doubler is illustrated
in figure 19. When high transconductance tubes
are employed, it is necessary to employ a
split- stator grid tank capacitor to prevent self
oscillation; with well screened tetrodes or
pentodes having medium values of transconductance, a split -coil arrangement with a single- section capacitor may be employed (the
20
PUSH -PULL
FREQUENCY TRIPLER
The push -pull tripler is advantageous in the
v -h-f ronge since circuit balance is maintained both in the input and output circuits.
If the circuit is neutralized it may be used
either as a straight amplifier or as a tripler.
Either triodes or tetrodes may be used; dual unit tetrodes such as the 815, 832A, and
8298 are particularly effective in the v-h -f
range.
center tap of the grid coil being by- passed to
ground).
Push -Push
Multipliers
Push -Pull Frequency
It is frequently desirable
in the case of u -h -f and
v -h -f transmitters
that
frequency multiplication stages be balanced
with respect to ground. Further it is just as
easy in most cases to multiply the crystal or
v -f -o frequency by powers of three rather than
multiplying by powers of two as is frequently
done on lower frequency transmitters. Hence
the use of push -pull tripiers has become quite
prevalent in both commercial and amateur
v -h -f and u -h -f transmitter designs. Such stages
are balanced with respect to ground and appear
in construction and on paper essentially the
same as a push -pull r -f amplifier stage with
the exception that the output tank circuit is
tuned to three times the frequency of the grid
tank circuit. A circuit for a push -pull tripler
stage is shown in figure 20.
A push -pull tripler stage has the further
advantage in amateur work that it can also be
used as a conventional push -pull r -f amplifier
merely by changing the grid and plate coils
so that they tune to the same frequency. This
is of some advantage in the case of operating
the 50 -Mc. band with 50 -bic. excitation, and
then changing the plate coil to tune to 144
Mc. for operation of the stage as a tripler from
excitation on 48 Mc. This circuit arrangement
is excellent for operation with push -pull beam
tetrodes such as the 6360 and 829B, although
a pair of tubes such as the 2E26, or 5763 could
just as well be used if proper attention were
given to the matter of screen -lead inductance.
Tripiers
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Tank
HANDBOOK
Tank Circuit
Capacitances
13 -9
It is necessary that the proper value of Q
plate tank circuit of any r -f
amplifier. The following section has been devoted to a treatment of the subject, and charts
are given to assist the reader in the determination of the proper L/C ratio to be used in a
radio -frequency amplifier stage.
A Class C amplifier draws plate current in
the form of very distorted pulses of short duration. Such an amplifier is always operated into a tuned inductance- capacitance or tank circuit which tends to smooth out these pulses,
by its storage or tank action, into a sine wave
DYNAMIC
CHARACTERISTIC
Tank Circuit
Q
As stated before, the tank cir-
cuit of a Class C amplifier
receives energy in the form of short pulses of
plate current which flow in the amplifier tube.
But the tank circuit must be able to store
enough energy so that it can deliver a current
essentially sine wave in form to the load. The
ability of a tank to store energy in this manner may be designated as the effective Q of
the tank circuit. The effective circuit Q may
be stated in any of several ways, but essentially the Q of a tank circuit is the ratio of the
energy stored to 2e times the energy lost per
cycle. Further, the energy lost per cycle must,
by definition, be equal to the energy delivered
to the tank circuit by the Class C amplifier
tube or tubes.
The Q of a tank circuit at resonance is equal
to its parallel resonant impedance (the resonant impedance is resistive at resonance) divided by the reactance of either the capacitor or the inductor which go to make up the
tank. The inductive reactance is equal to the
capacitive reactance, by definition, at resonance. Hence we may state:
259
A_
(0\
be used in the
of radio -frequency output. Any wave -form distortion of the carrier frequency results in harmonic interference in higher- frequency channels.
A Class A r-f amplifier would produce a sine
wave of radio -frequency output if its exciting
waveform were also a sine wave. However, a
Class A amplifier stage converts its d -c input
to r -f output by acting as a variable resistance,
and therefore heats considerably. A Class C
amplifier when driven hard with short pulses
at the peak of the exciting waveform acts more
as an electronic switch, and therefore can convert its d -c input to r -f output with relatively
good efficiency. Values of plate circuit efficiency from 65 to 85 per cent are common in
Class C amplifiers operating under optimum
conditions of excitation, grid bias, and loading.
Circuits
GRID SWING
Figure
21
AMPLIFIER OPERATION
Plate current pulses are shown at (A), (e),
and (C). The dip in the top of the plate current waveform will occur when the excitation
voltage is such that the minimum plate voltage dips below the maximum grid voltage.
A detailed discussion of the operation of
Class C amplifiers is given in Chapter Seven.
CLASS
C
Q =
-=RL
RL
Xc
XL
where RL is the resonant impedance of the
tank and Xc is the reactance of the tank capacitor and XL is the reactance of the tank
coil. This value of resonant impedance, RL,
is the load which is presented to the Class C
amplifier tube in a single -ended circuit such
as shown in figure 21.
The value of load impedance, RL, which the
Class C amplifier tube sees may be obtained,
looking in the other direction from the tank
coil, from a knowledge of the operating conditions on the Class C tube. This load impedance may be obtained from the following expression, which is true in the general case of
any Class C amplifier:
Epm=
RL
2
Np lb Ebb
where the values in the equation have the characteristics listed in the beginningof Chapter 6.
The expression above is academic, since
the peak value of the fundamental component
of plate voltage swing, Epm, is not ordinarily
known unless a high -voltage peak a-c voltmeter
is available for checking. Also, the decimal
value of plate circuit efficiency is not ordinarily known with any degree of accuracy. However, in a normally operated Class C amplifier
Generation of
260
R
> s
w
>
!
54
V-
8 3.
u x
100
,S
10
TANK CIRCUIT
30
25
20
Q
THE
Energy
-F
which means simply that the resistance presented by the tank circuit to the Class C tube
is approximately equal to one -half the d -c load
resistance which the Class C stage presents
to the power supply (and also to the modulator
in case high -level modulation of the stage is
to be used).
Combining the above simplified expression
for the r -f impedance presented by the tank to
the tube, with the expression for tank Q given
in a previous paragraph we have the following
expression which relates the reactance of the
tank capacitor or coil to the d-c input to the
Class
C
stage:
XC
Figure 22
RELATIVE HARMONIC OUTPUT
PLOTTED AGAINST TANK CIRCUIT
.v
Rd. c.
\\\\1
\
\
II
2
20000
15
10
i
Ó
>
w
I-
XL
,
Rd.c.
The above expression is the basis of the
usual charts giving tank capacitance for the
various bands in terms of the d -c plate voltage
and current to the Class C stage, including
the charts of figure 23, figure 24 and figure 25.
Harmonic Rodio-
The problem of
harmonic
radiation from transmitters
has long been present, but
it has become critical only relatively recently
along with the extensive occupation of the
v -h -f range. Television signals are particularly
susceptible to interference from other signals
falling within the pass band of the receiver,
so that the TVI problem has received the major
emphasis of all the services in the v -h -f range
which are susceptible to interference from
harmonics of signals in the h -f or lower v -h -f
range.
tion vs.
Q
\11111III1N1
111111!\IIIIIII
\\
Q=12
II
w
=
2Q
Q
the plate voltage swing will be approximately
equal to 0.85 to 0.9 times the d -c plate voltage
on the stage, and the plate circuit efficiency
will be from 70 to 80 per cent (Np of 0.7 to
0.8), the higher values of efficiency normally
being associated with the higher values of
plate voltage swing. With these two assumptions as to the normal Class C amplifier, the
expression for the plate load impedance can
be greatly simplified to the following approximate but useful expression:
RL
RADIO
III
IIII
mum
111
.
11
1
NEUTRALIZING
COIL
\11'I
\\111Ii
1111, I\IIII
(I
RFC
-e
1110M1121111101111111111
3
10
20
IIII!ÍiNHO!i
30
100
500
200
TOTAL CAPACITANCE ACROSS LC
C
1000
2000
RCUIT (CO
O
Figure 23
PLATE -TANK CIRCUIT ARRANGEMENTS
Shown above in the case of each of the tank circuit types is the recommended tank circuit capacitance. (A) is a conventional tetrode amplifier, (B) is a coil -neutralized triode amplifier,
(C) is a grounded-grid triode amplifier, (D) is a grid -neutralized triode amplifier.
www.americanradiohistory.com
re
HANDBOOK
10
Tank
Circuits
261
\.,'..O..U...n
\/
\11 III
IIII
ÌIÌI!I\1iIÌCIÌI\\IIIIIÌ
MONIIIMMIIIMUM111
\\111i111\11111111111111
1MINE111011; I111111MII1,vIM11111MMnII
1.3
o
>
á
II
¢
1
11
1
1
41 1
11
1
1
.,
IlliliE1111P11 1111111
et
11111111111111111111
10
2 3
5 7
lo 30 50 100 lao
500 1000
CORRECT VALUES OF TANK CIRCUIT CAPACITANCE (C FOR
OPERATING Q OF 12 WITH SINGLE-ENDED SPLIT TANK COILS
Figure 24
PLATE -TANK CIRCUIT ARRANGEMENTS
Shown above for each of the tank circuit types is the recommended tank circuit capacitance of
the operating frequency for an operating Q of 12. (A) is a split -stator tank, each section of which
is twice the capacity value read on the graph. (8) is circuit using tapped coil for phase reversal.
Inspection of figure 22 will show quickly
that the tank circuit of a Class C amplifier
should have an operating Q of 12 or greater
to afford satisfactory rejection of second harmonic energy. The curve begins to straighten
out above a Q of about 15, so that a considerable increase in Q must be made before an appreciable reduction in second -harmonic energy
is obtained. Above a circuit Q of about 10 any
increase will not afford appreciable reduction
in the third -harmonic energy, so that additional
harmonic filtering circuits external to the amplifier proper must be used if increased attenuation of higher order harmonics is desired.
The curves also show that push -pull amplifiers
may be operated at Q values of 6 or so, since
the second harmonic is cancelled to a large
extent if there is no unbalanced coupling between the output tank circuit and the antenna
system.
Figures 23, 24 and 25 illustrate the correct value
of tank capacity for various circuit configurations. A Q value of 12
has been chosen as optimum for single ended
circuits, and a value of 6 has been chosen for
push -pull circuits. Figure 23 is used when a
single ended stage is employed, and the capacitance values given are for the total capacitance across the tank coil. This value includes the tube interelectrode capacitance
(plate to ground), coil distributed capacitance,
wiring capacities, and the value of any lowCapacity Charts for
Correct Tank
Q
inductance plate -to- ground by -pass capacitor
as used for reducing harmonic generation, in
addition to the actual "in -use" capacitance
of the plate tuning capacitor. Total circuit
stray capacitance may vary from perhaps 5
micromicrofarads for a v -h -f stage to 30 micro microfarads for a medium power tetrode h -f
stage.
When a split plate tank coil is employed in
the stage in question, the graph of figure 24
should be used. The capacity read from the
graph is the total capacity across the tank
coil. If the split- stator tuning capacitor is
used, each section of the capacitor should
have a value of capacity equal to twice the
value indicated by the graph. As in the case
of figure 23, the values of capacity read on
the graph of figure 24 include all residual circuit capacities.
For push-pull operation, the correct values
of tank circuit capacity may be determined
with the aid of figure 25. The capacity values
obtained from figure 25 are the effective values
across the tank circuit, and if a split- stator
tuning capacitor is used, each section of the
capacitor should have a value of capacity equal to twice the value indicated by the graph.
As in the case of figures 23 and 24, the values
of capacity read on the graph of figure 25 include all residual circuit capacities.
The tank circuit operates in the same manner whether the tube feeding it is a pentode,
beam tetrode, neutralized triode, grounded grid triode, whether it is single ended or push-
www.americanradiohistory.com
Generation of
262
R
-F
THE
Energy
RADIO
20000
Q°6
16000
0
8000
uz
á
>ú
«
6000
0:
w
J
O. a.
ú
é
1000
2
3
!
7
10
20
30
30
00
200
300 1000
) FOR
CORRECT VALUES OF TANK CIRCUIT CAPACITANCE C
OPERAT NG Q OF 6 WITH PUSH -PULL TANK CIRCUITS
Figure 25
PLATE -TANK CIRCUIT ARRANGEMENTS FOR PUSH -PULL STAGES
Shown above is recommended tank circuit capacity at operating frequency for a Q of 6. (A) is
split-stator tank, each section of which is twice the capacity value read on the graph. (B) is
circuit using topped coil for phase reversal.
pull, or whether it is shunt fed or series fed.
The important thing in establishing the operating Q of the tank circuit is the ratio of the
loaded resonant impedance across its terminals to the reactance of the L and the C which
make up the tank.
Due to the unknowns involved in determining circuit stray capacitances it is sometimes
more convenient to determine the value of L
required for the proper circuit Q (by the method
discussed earlier in this Section) and then to
vary the tuned circuit capacitance until resonance is reached. This method is most frequently used in obtaining proper circuit Q in
commercial transmitters.
The values of Rp for using the charts are
easily calculated by dividing the d -c plate supply voltage by the total d -c plate current (expressed in amperes). Correct values of total
tuning capacitance are shown in the chart for
the different amateur bands. The shunt stray
capacitance can be estimated closely enough
for all practical purposes. The coil inductance
should then be chosen which will produce
resonance at the desired frequency with the
total calculated tuning capacitance.
The Q of a circuit depends
upon the resistance in series
with the capacitance and inductance. This series resistance is very low
for a low -loss coil not loaded by an antenna
circuit. The value of Q may be from 100 to 600
under these conditions. Coupling an antenna
Effect of Loading on
Q
circuit has the effect of increasing the series
resistance, though in this case the power is
consumed as useful radiation by the antenna.
Mathematically, the antenna increases the
value of R in the expression Q = oiL /R where
L is the coil inductance in microhenrys and
is the term 2nf, f being in megacycles.
The coupling from the final tank circuit to
the antenna or antenna transmission line can
be varied to obtain values of Q from perhaps
3 at maximum coupling to a value of Q equal
to the unloaded Q of the circuit at zero antenna coupling. This value of unloaded Q can
be as high as 500 or 600, as mentioned in the
preceding paragraph. However, the value of
Q = 12 will not be obtained at values of normal d -c plate current in the Class C amplifier
stage unless the C -to -L ratio in the tank circuit is correct for that frequency of operation.
To determine the required
tuning capacitor air gap for
a particular amplifier circuit it is first necessary to estimate the peak
r-f voltage which will appear between the
plates of the tuning capacitor. Then, using
figure 26, it is possible to estimate the plate
spacing which will be required.
The instantaneous r -f voltage in the plate
circuit of a Class C amplifier tube varies from
nearly zero to nearly twice the d -c plate voltage. If the d -c voltage is being 100 per cent
modulated by an audio voltage, the r-f peaks
will reach nearly four times the d -c voltage.
Tuning Capacitor
Air Gap
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HANDBOOK
L
and
Pi
Networks
RP RA(Q2+1)(txACT)
FIGURE 26
USUAL BREAKDOWN RATINGS OF
COMMON PLATE SPACINGS
Air-gap in
Peak voltage
inches
breakdown
.030
1,000
.050
2,000
.070
3,000
.100
4,000
.125
4,500
.150
5,200
.170
6,000
.200
7,500
.250
9,000
.350
11,000
.500
15,000
.700
20,000
Recommended air -gap for use when no d -c
voltage appears across plate tank condenser
(when plate circuit is shunt fed, or when the
plate tank condenser is insulated from
ground).
D.C. PLATE
VOLTAGE
400
600
750
1000
1250
1500
2000
2500
3000
3500
263
C.W.
PLATE
MOD.
.030
.050
.050
.070
.050
.070
.084
.100
.070
RP
=
Q2 RA (APPROX.)
Q=xs__X.. -BL-B.e
RA
RA
XC
XL
XL =Xc
RF
+e
RP= APPROX. LATE VOLTAG4
'PLATE CURRENT
RP= 225 RA
=
FOR OPERATING
Q OF 15
CIRCUIT
XC
XL=
Figure
THE
*
19
27
NETWORK
IMPEDANCE
TRANSFORMER
The L network is useful with a moderate
operating Q for high values of impedance
transformation, and it may be used for applications other than in the plate circuit of a
tube with relatively low values of operating
Q for moderate impedance transformations.
Exact and approximate design equations ore
given.
L
.144
.078
.100
.175
.200
.250
.200
.250
.375
.500
.600
should be multiplied by 1.5 for
some safety factor when d -c voltage appears
Spacings
across plate tank condenser.
These rules apply to a loaded amplifier or
buffer stage. If either is operated without an
r -f load, the peak voltages will be greater and
can exceed the d-c plate supply voltage. For
this reason no amplifier should be operated
without load when anywhere near normal d -c
plate voltage is applied.
If a plate blocking condenser is used, it
must be rated to withstand the d -c plate voltage plus any audio voltage. This capacitor
should be rated at a d -c working voltage of at
least twice the d-c plate supply in a plate modulated amplifier, and at least equal to the d -c
supply in any other type of r -f amplifier.
between the plate tank circuit of an amplifier
and a transmission line, or they may be used
to match directly from the plate circuit of an
amplifier to the line without the requirement
for a tank circuit -provided the network is designed in such a manner that it has sufficient
operating Q for accomplishing harmonic attenuation.
The L Matching
Network
L and Pi Matching
Networks
The L network is of limited
utility in impedance matching since its ratio of impedance transformation is fixed at a value equal
to (Q2 +1). The operating Q may be relatively
low (perhaps 3 to 6) in a matching network between the plate tank circuit of an amplifier
and a transmission line; hence impedance
transformation ratios of 10 to 1 and even lower
may be attained. But when the network also
acts as the plate tank circuit of the amplifier
stage, as in figure 27, the operating Q should
be at least 12 and preferably 15. An operating
Q of 15 represents an impedance transformation of 225; this value normally will be too
high even for transforming from the 2000 to
10,000 ohm plate impedance of a Class C amplifier stage down to a 50 -ohm transmission
The L and pi networks often can be put to
advantageous use in accomplishing an impedance match between two differing impedances.
Common applications are the matching between
a transmission line and an antenna, or between
the plate circuit of a single -ended amplifier
stage and an antenna transmission line. Such
networks may be used to accomplish a match
However, the L network is interesting since
the basis of design for the pi network.
Inspection of figure 27 will show that the L
network in reality must be considered as a
parallel- resonant tank circuit in which RA
represents the coupled -in load resistance;
only in this case the load resistance is directly coupled into the tank circuit rather than
being inductively coupled as in the conven-
13 -10
line.
it forms
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Generation of
264
R
-F
THE
Energy
RADIO
pacitance may be obtained for an operating Q
of 12 by reference to figures 23, 24 and 25.
The inductive arm in the pi network can be
thought of as consisting of two inductances
in series, as illustrated in figure 28. The first
is that value of
portion of this inductance,
inductance which would resonate with C, at
the operating frequency -the same as in a conventional tank circuit. However, the actual
value of inductance in this arm of the pi network, L10, will be greater than L, for normal
values of impedance transformation. For high
transformation ratios Lot will be only slightly
greater than Li; for a transformation ratio of
1.0, L10t will be twice as great as L. The
amount of inductance which must be added to
L, to restore resonance and maintain circuit
Q is obtained through use of the expression
L
Roc
Ebb
=
XCZ
I
Rp
XCn
-RA
Rp
RA(Q2+1)-Rp
Roc.
z
Rp
=
XCi
RI12tXG22
XLZ'
RAZ
ALTOT.
XL1+XL2
Q
XL,
Á
for X12 in figure 28.
Figure 28
THE PI NETWORK
The pi network is valuable for use as an impedance transformer over a wide ratio of
transformation values. The operating Q should
be at least 12 and preferably 15 to 20 when
the circuit is to be used in the plate circuit
of
Class C amplifier. Design equations
are given above. The inductor Ltot represents a single inductance, usually variable,
with a value equal to the sum of Lt and L2.
a
tional arrangement where the load circuit is
coupled to the tank circuit by means of a link.
When RA is shorted, L and C comprise a conventional parallel- resonant tank circuit, since
for proper operation L and C must be resonant
in order for the network to present a resistive
load to the Class C amplifier.
pi impedance matching
network, illustrated in figure
28, is much more general in its application
than the L network since it offers greater harmonic attenuation, and since it can be used
to match a relatively wide range of impedances
while still maintaining any desired operating
Q. The values of C, and L, in the pi network
of figure 28 can be thought of as having the
same values of the L network in figure 27 for
the same operating Q, but what is more important from the comparison standpoint these values will be the same as in a conventional tank
circuit.
The value of the capacitance may be determined by calculation, with the operating Q and
the load impedance which should be reflected
to the plate of the Class C amplifier as the
two knowns -or the actual values of the ca-
The Pi Network
The
The peak voltage rating of the main tuning
capacitor C, should be the normal value for a
Class C amplifier operating at the plate volt-
age to be employed. The inductor L101 may be
a plug -in coil which is changed for each band
of operation, or some sort of variable inductor
may be used. A continuously variable
slider -
type of variable inductor, such as used in certain items of surplus military equipment, may
be used to good advantage if available, or a
tapped inductor such as used in the ART -13
may be employed. However, to maintain good
circuit Q on the higher frequencies when a
variable or tapped coil is used on the lower
frequencies, the tapped or variable coil should
be removed from the circuit and replaced by
a smaller coil which has been especially designed for the higher frequency ranges.
The peak voltage rating of the output or
loading capacitor, C2i is determined by the
power level and the impedance to be fed. If a
50 -ohm coaxial line is to be fed from the pi
network, receiving -type capacitors will be
satisfactory even up to the power level of a
plate -modulated kilowatt amplifier. In any
event, the peak voltage which will be impressed across the output capacitor is expressed by: Epk2 = 2 R. Wo, where Epk is the
peak voltage across the capacitor, R. is the
value of resistive load which the network is
feeding, and W. is the maximum value of the
average power output of the stage. The harmonic attenuation of the pi network is quite
good, although an external low -pass filter will
be required to obtain harmonic attenuation
value upward of 100 db such as normally required. The attenuation to second harmonic
energy will be approximately 40 db for an operating Q of 15 for the pi network; the value
increases to about 45 db for a 1:1 transformation and falls to about 38 db for an impedance
step -down of 80:1, assuming that the operating Q is maintained at 15.
www.americanradiohistory.com
HANDBOOK
Grid
Bias
265
RFC
CB
COAX
OUTPUT
-B
E.
Zx[B
PLATE LOAD (OHMS)
WHERE ES IS PLATE VOLTAGE
AND I B IS PLATE CURRENT
IN
4MPE II
CS
Ce- .000252/F
MICA CAPACITOR RATED AT TWICE THE D.C.
PLATE VOLTAGE
RFC i -Ne 28 ENAMELED. CLOSE -WOUND ON
I'OIA.,
RFC2Est
mated
I.
n
1,500
2,000
2,500
3,000
3,500
4,000
4,500
5,000
7
14
21
360
180
90
60
210
105
180
90
120
60
110
56
52
35
45
34
23
30
20
28
65
45
26
33
155
76
38
25
19
135
68
28
280
140
70
47
35
17
15
4.5
2.2
3.5 Mc
in puf, 3.5 Mc
7
14
21
28
gal,
6.5
3.2
1.6
3.5 Mc
7
14
21
28
31
8.5
4.2
10.5
5.2
12.5
6.2
14
15.5
7
2.6
3.1
7.8
3.9
2.6
4.5
1.95
2.25
18
9
19
14
20
10
5
6,000
NOTES
actuel capacitance setting
45 for C, equals the value in this
23 table minus the published tube
15 output
capacitance. Air gap
11
approx. 10 mils 100 v E,,.
90
The
25
Inductance
values
are for
a
12.5 50 -ohm load. For a 70 -ohm
6.2 load, values ore approx. 3%
0.73
0.55
1.08
2.1
1.38
2.05
0.8
1.05
1.7
1.28
3.5
2.3
1.55
1.7
2,400
1,200
600
400
300
2,100
1,060
530
350
265
1,800
900
1,550
760
1,400
700
1,250
630
1,100
560
1,000
500
900
460
450
300
225
380
250
190
350
230
175
320
210
160
280
250
230
700 For 50 -ohm transmission line.
350 Air gap for Cr is approx. 1
175 mil 100 v E,.
185
140
165
125
155
115
120
90
1.800
900
450
300
225
1,500
750
370
250
185
1,300
650
320
215
160
1,100
560
280
1,000
500
250
900
450
220
800
400
200
720
360
180
640
320
160
190
170
125
145
110
130
100
120
110
90
80
500
250
125
85
65
1.1
28
Cr in
2,1 MN, NATIONAL R-100
520
260
130
85
7
14
21
C1
CERAMIC INSULATOR
1,000
µµf, 3.5 Mc
in ph,
A
R-1754
Plate
load (ohms)
C,
4 -LONG OR NATIONAL
140
3
3.3
2.5
4.1
3.1
higher.
For
70 -ohm
transmission line.
are for a Q of 12. for other values of Q, use
Values given are approximations. All components shown in Table
L,.
Q.
C.
Q.
is
higher
than 5,000 ohms, it is recommended that the
When the estimated plate load
and _
1
Q,,
-C,,
Q
L.
components be selected for a circuit Q between 20 and 30.
Table
1
Components for Pi- Coupled Final Amplifiers
To simplify design pro cedure, a pi- network chart,
compiled by M. Seybold,
W2RYI (reproduced by courtesy of R.C.A. Tube
Division, Harrison, N.J.) is shown in table 1.
This chart summarizes the calculations of figure 28 for various values of plate load.
Component Chart
for Pi- Networks
13 -11
Grid Bias
Radio-frequency amplifiers require some
form of grid bias for proper operation. Practically all r -f amplifiers operate in such a manner that plate current flows in the forni of
short pulses which have a duration of only a
fraction of an r -f cycle. To accomplish this
with a sinusoidal excitation voltage, the operating grid bias must be at least sufficient to
cut off the plate current. In very high efficiency Class C amplifiers the operating bias may
be many times the cutoff value. Cutoff bias,
it will be recalled, is that value of grid voltage which will reduce the plate current to
zero at the plate voltage employed. The method
for calculating it has been indicated previously. This theoretical value of cutoff will not
reduce the plate current completely to zero,
due to the variable-ft tendency or "knee"
which is characteristic of all tubes as the
cutoff point is approached.
Amplitude modulated Class C
amplifiers should be operated
with the grid bias adjusted to a value greater
than twice cutoff at the operating plate voltClass
C
Bias
Generation of
266
FOU
R-
F
THE
Energy
RADIO
FROLIC/RIVER
DRIVER
Figure 29
GRID -LEAK BIAS
The grid leak on an amplifier or multiplier
stage may also be used as the shunt feed
impedance to the grid of the tube when o
high value of grid leak (greater than perhaps
20,000 ohms) is used. When a lower value of
grid leak is to be employed, an r -f choke
should be used between the grid of the tube
and the grid leak to reduce r -f losses in the
grid leak resistance.
age. This procedure will insure that the tube
is operating at a bias greater than cutoff when
the plate voltage is doubled on positive modulation peaks. C -w telegraph and FM transmitters can be operated with bias as low as
cutoff, if only limited excitation is available
and moderate plate efficiency is satisfactory.
In a c -w transmitter, the bias supply or resistor should be adjusted to the point which
will allow normal grid current to flow for the
particular amount of grid driving r -f power
available. This form of adjustment will allow
more output from the under -excited r -f amplifier than when higher bias is used with corresponding lower values of grid current. In any
event, the operating bias should be set at as
low a value as will give satisfactory operation, since harmonic generation in a stage increases rapidly as the bias is increased.
resistor can be connected
Class
C amplifier to provide grid -leak bias. This resistor, R, in figure 29, is part of the d -c path
Grid -Leak Bias
A
in the grid circuit of a
circuit.
The r -f excitation applied to the grid circuit of the tube causes a pulsating direct current to flow through the bias supply lead, due
to the rectifying action of the grid, and any
current flowing through R, produces a voltage
drop across that resistor. The grid of the tube
is positive for a short duration of each r -f
cycle, and draws electrons from the filament
or cathode of the tube during that time. These
electrons complete the circuit through the d -c
grid return. The voltage drop across the resistance in the grid return provides a negain the grid
tive bias for the grid.
Grid-leak bias automatically adjusts itself
over fairly wide variations of r -f excitation.
The value of grid-leak resistance should be
such that normal values of grid current will
flow at the maximum available amount of r -f
Figure 30
COMBINATION GRID -LEAK AND
FIXED BIAS
Grid -leak bias often is used in conjunction
with a fixed minimum value of power supply
bias. This arrangement permits the operating
bias to be established by the excitation energy, but in the absence of excitation the electrode currents to the tube will be held to safe
values by the fixed- minimum power supply
bias. If a relatively low value of grid leak
is to be used, an r -f choke should be connected between the grid of the tube and the
grid leak as discussed in figure 29.
excitation. Grid -leak bias cannot be used for
grid -modulated or linear amplifiers in which
the average d -c grid current is constantly
varying with modulation.
Safety Bias
Grid -leak bias alone provides no
protection against e x c e s s i v e
plate current in case of failure of the source
of r -f grid excitation. A C- battery or C -bias
supply can be connected in series with the
grid leak, as shown in figure 30. This fixed
protective bias will protect the tube in the
event of failure of grid excitation. "Zero- bias"
tubes do not require this bias source in addition to the grid leak, since their plate current
will drop to a safe value when the excitation
is removed.
resistor can be connected in
series with the cathode or center- tapped filament lead of an amplifier to secure automatic bias. The plate current flows
through this resistor, then back to the cathode
or filament, and the voltage drop across the
resistor can be applied to the grid circuit by
connecting the grid bias lead to the grounded
or power supply end of the resistor R, as shown
Cathode Bias
A
in figure 31.
The grounded (B- minus) end of the cathode
resistor is negative relative to the cathode
by an amount equal to the voltage drop across
the resistor. The value of resistance must be
so chosen that the sum of the desired grid
and plate current flowing through the resistor
will bias the tube for proper operation.
This type of bias is used more extensively
in audio-frequency than in radio -frequency amplifiers. The voltage drop across the resistor
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Protective Circuits
HANDBOOK
267
PROM
DRIVER
Figure
Figure
31
R
RF STAGE WITH CATHODE BIAS
Cathode bias sometimes is advantageous for
use it on r -f stage that operates with a relatively small amount of r -f excitation.
must be subtracted from the total plate supply
voltage when calculating the power input to
the amplifier, and this loss of plate voltage
in an r-f amplifier may be excessive. A Class
A audio amplifier is biased only to approximately one -half cutoff, whereas an r -f amplifier
may be biased to twice cutoff, or more, and
thus the plate supply voltage loss may be a
large percentage of the total available voltage
when using low or medium It tubes.
Oftentimes just enough cathode bias is employed in an r -f amplifier to act as safety bias
to protect the tubes in case of excitation failure, with the rest of the bias coming from a
grid leak.
Separate Bias
Supply
An external supply often is
used for grid bias, as shown in
figure 32. Battery bias gives
very good voltage regulation and is satisfactory for grid- modulated or linear amplifiers,
which operate at low grid current. In the case
of Class C amplifiers which operate with high
grid current, battery bias is not satisfactory.
This direct current has a charging effect on
the dry batteries; after a few months of service
the cells will become unstable, bloated, and
noisy.
A separate
a -c operated power supply is
commonly used for grid bias. The bleeder resistance across the output of the filter can be
made sufficiently low in value that the grid
current of the amplifier will not appreciably
change the amount of negative grid -bias voltage. Alternately, a voltage regulated grid -bias
supply can be used. This type of bias supply
is used in Class B audio and Class B r-f linear amplifier service where the voltage regulation in the C-bias supply is important. For
a Class C amplifier, regulation is not so important, and an economical design of components in the power supply, therefore, can be
utilized. In this case, the bias voltage must
be adjusted with normal grid current flowing,
as the grid current will raise the bias con-
-F
32
STAGE WITH BATTERY
BIAS
Battery bias is seldom used, due to deterioration of the cells by the reverse grid current.
However, it may be used in certain special
applications, or the fixed bias voltage may
be supplied by a bias power supply.
siderably when it is flowing through the bias supply bleeder resistance.
13 -12
Protective Circuits for
Tetrode Transmitting Tubes
The tetrode transmitting tube requires three
operating voltages: grid bias, screen voltage,
and plate voltage. The current requirements of
these three operating voltages are somewhat
interdependent, and a change in potential of
one voltage will affect the current drain of the
tetrode in respect to the other two voltages.
In particular, if the grid excitation voltage is
interrupted as by keying action, or if the plate
supply is momentarily interrupted, the resulting
voltage or current surges in the screen circuit
are apt to permanently damage the tube.
simple method of obtaining screen voltage is by
means of a dropping resistor from the high voltage plate supply, as shown
in figure 33. Since the current drawn by the
screen is a function of the exciting voltage
applied to the tetrode, the screen voltage will
rise to equal the plate voltage under conditions of no exciting voltage. If the control grid
is overdriven, on the other hand, the screen
current may become excessive. In either case,
damage to the screen and its associated components may result. In addition, fluctuations
in the plate loading of the tetrode stage will
cause changes in the screen current of the
tube. This will result in screen voltage fluctuations due to the inherently poor voltage
regulation of the screen series dropping resistor. These effects become dangerous to tube
life if the plate voltage is greater than the
screen voltage by a factor of 2 or so.
The Series Screen
Supply
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A
268
Generation of
R
-F
THE
Energy
RADIO
RFC
r
NEGATIVE
OPERATING
Figure 33
8/AS CUTS
OFF
DROPPING- RESISTOR SCREEN SUPPLY
The Clomp Tube
CLAMP{
rUBE
4B
CLAMP
TUBE
Figure 34
CLAMP -TUBE SCREEN SUPPLY
A clamp tube may be added
to the series screen supply,
as shown in figure 34. The clamp tube is normally cut off by virtue of the d -c grid bias drop
developed across the grid resistor of the tetrode tube. When excitation is removed from
the tetrode, no bias appears across the grid
resistor, and the clamp tube conducts heavily,
dropping the screen voltage to a safe value.
When excitation is applied to the tetrode the
clamp tube is inoperative, and fluctuations of
the plate loading of the tetrode
allow the screen voltage to rise to
value. Because of this factor, the
does not offer complete protection
rode.
tube could
a damaging
clamp tube
to the tet-
A low voltage
may be used
screen supply
of the
series screen dropping resistor. This will protect the screen circuit from
excessive voltages when the other tetrode
operating parameters shift. However, the screen
can be easily damaged if plate or bias voltage is removed from the tetrode, as the screen
current will reach high values and the screen
dissipation will be exceeded. If the screen
supply is capable of providing slightly more
screen voltage than the tetrode requires for
proper operation, a series wattage -limiting resistor may be added to the circuit as shown
in figure 35. With this resistor in the circuit
it is possible to apply excitation to the tetrode tube with screen voltage present (but in
the absence of plate voltage) and still not damage the screen of the tube. The value of the
resistor should be chosen so that the product
of the voltage applied to the screen of the
tetrode times the screen current never exceeds
the maximum rated screen dissipation of the
tube.
The Separate
Screen Supply
instead
piing. The latter is a special form of inductive coupling. The choice of a coupling method
depends upon the purpose for which it is to
be used.
Capacitive coupling between an
amplifier or doubler circuit and a
preceding driver stage is shown
in figure 36. The coupling capacitor, C, isolates the d -c plate supply from the next grid
and provides a low impedance path for the r-f
energy between the tube being driven and the
driver tube. This method of coupling is simple
and economical for low power. amplifier or exciter stages, but has certain disadvantages,
particularly for high frequency stages. The
grid leads in an amplifier should be as short
as possible, but this is difficult to attain in
the physical arrangement of a high power amplifier with respect to a capacitively- coupled
driver stage.
Capacitive
Coupling
One significant disadvantage of capacitive coupling
is the difficulty of adjusting
the load on the driver stage.
Impedance adjustment can be accomplished
by tapping the coupling lead a part of the way
down on the plate coil of the tuned stage of
the driver circuit; but often when this is done
Disadvantages of
Capacitive
Coupling
SERIES RESISTOR
LOW VOLTAGE
SCREEN SUPPLY
13 -13
+B
Interstage Coupling
Energy is usually coupled from one circuit
of a transmitter into another either by capacitive coupling, inductive coupling, or link cou-
Figure 35
PROTECTIVE WATTAGE -LIMITING RESISTOR FOR USE WITH LOW- VOLTAGE
SCREEN SUPPLY
A
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HANDBOOK
Figure 36
CAPACITIVE INTERSTAGE COUPLING
parasitic oscillation will take place in the
stage being driven.
One main disadvantage of capacitive coupling lies in the fact that the grid -to- filament
capacitance of the driven tube is placed directly across the driver tuned circuit. This
condition sometimes makes the r -f amplifier
difficult to neutralize, and the increased minimum circuit capacitance makes it difficult to
use a reasonable size coil in the v -h -f range.
Difficulties from this source can be partially
eliminated by using a center-tapped or split stator tank circuit in the plate of the driver
stage, and coupling capacitively to the opposite end from the plate. This method places
the plate -to- filament capacitance of the driver
across one -half of the tank and the grid -tofilament capacitance of the following stage
across the other half. This type of coupling is
a
shown in figure 37.
Capacitive coupling can be used to advantage in reducing the total number of tuned circuits in a transmitter so as to conserve space
and cost. It also can be used to advantage between stages for driving beam tetrode or pentode amplifier or doubler stages.
Inductive coupling (figure 38) results when two coils are electromagnetically coupled to one another. The degree of coupling is controlled by
varying the mutual inductance of the two coils,
which is accomplished by changing the spacing or the relationship between the axes of
the coils.
Inductive
Coupling
Interstage
Coupling
269
Figure 37
BALANCED CAPACITIVE COUPLING
Balanced capacitive coupling sometimes is
useful when it is desirable to use o relatively
large inductance in the interstage tank circuit, or where the exciting stage is neutralized as shown above.
Inductive coupling is used extensively for
coupling r -f amplifiers in radio receivers. However, the mechanical problems involved in adjusting the degree of coupling limit the usefulness of direct inductive coupling in transmitters. Either the primary or the secondary
or both coils may be tuned.
If the grid tuning capacitor of
figure 38 is removed and the
coupling increased to the maximum practicable
Unity Coupling
value by interwinding the turns of the two coils,
the circuit insofar as r.f. is concerned acts
like that of figure 36, in which one tank serves
both as plate tank for the driver and grid tank
for the driven stage. The inter-wound grid
winding serves simply to isolate the d-c plate
voltage of the driver from the grid of the driven
stage, and to provide a return for d -c grid current. This type of coupling, illustrated in figure 39, is commonly known as unity coupling.
Because of the high mutual inductance, both
primary and secondary are resonated by the
one tuning capacitor.
INTERWOUND
Figure 39
"UNITY" INDUCTIVE COUPLING
Figure 38
INDUCTIVE INTERSTAGE COUPLING
Due to the high value of coupling between
the two coils, one tuning capacitor tunes
both circuits. This arrangement often is usa
ful in coupling from a single -ended to a pushpull stage.
270
Generation of
R
-F
Energy
THE
LINK COUPLING
LINK COUPLING
AT
AT ..COLD. ENDS.
UPPER ENDS "MOT"
Figure 40
INTERSTAGE COUPLING BY MEANS
OF A
used since the two stages may be separated
by a considerable distance, since the amount
of a coupling between the two stages may he
easily varied, and since the capacitances of
the two stages may be isolated to permit use
of larger inductances in the v-h -f range.
special form of inductive
coupling which is widely employed in radio transmitter circuits is known
as link coupling. A low impedance r -f transmission line couples the two tuned circuits
together. Each end of the line is terminated
in one or more turns of wire, or links, wound
around the coils which are being coupled together. These links should be coupled to each
tuned circuit at the point of zero r -f potential,
or nodal point. A ground connection to one
side of the link usually is used to reduce harmonic coupling, or where capacitive coupling
between two circuits must be minimized. Coaxial line is commonly used to transfer energy
between the two coupling links, although Twin Lead may be used where harmonic attenuation
is not so important.
Typical link coupled circuits are shown in
figures 40 and 41. Some of the advantages of
link coupling are the following:
(1)
(2)
(3)
(4)
(5)
(6)
A
It eliminates coupling taps on tuned cir-
cuits.
It permits the use of series power supply
connections in both tuned grid and tuned
plate circuits, and thereby eliminates the
need of shunt -feed r -f chokes.
It allows considerable separation between
transmitter stages without appreciable
r-f losses or stray chassis currents.
It reduces capacitive coupling and thereby makes neutralization more easily attainable in r -f amplifiers.
It provides semi- automatic impedance
matching between plate and grid tuned
circuits, with the result that greater grid
drive can be obtained in comparison to
capacitive coupling.
It effectively reduces the coupling of harmonic energy.
COLO CENTER
ENDS "HOT*
Figure
"LINK"
PUSH -PULL
Link interstoge coupling is very commonly
Link Coupling
RADIO
41
LINK COUPLING
The link -coupling line and links can be
made of no. 18 push -back wire for coupling
between low -power stages. For coupling between higher powered stages the 150 -ohm
Twin -Lead transmission line is quite effective
and has very low loss. Coaxial transmission is
most satisfactory between high powered amplifier stages, and should always be used
where harmonic attenuation is important.
13 -14
Radio- Frequency
Chokes
Radio -frequency
chokes are connected in
circuits for the purpose of stopping the passage of r -f energy while still permitting a direct current or audio -frequency current to pass.
They consist of inductances wound with a
large number of turns, either in the form of a
solenoid, a series of solenoids, a single universal pie winding, or a series of pie windings. These inductors are designed to have as
much inductance and as little distributed or
shunt capacitance as possible. The unavoidable small amount of distributed capacitance
resonates the inductance, and this frequency
normally should be much lower than the frequency at which the transmitter or receiver
circuit is operating. R -f chokes for operation
on several bands must be designed carefully
so that the impedance of the choke will be extremely high (several hundred thousand ohms)
in each of the bands.
The direct current which flows through the
r -f choke largely determines the size of wire
to be used in the winding. The inductance of
r -f chokes for the v -h -f range is much less
than for chokes designed for broadcast and
ordinary short -wave operation. A very high
inductance r -f choke has more distributed capacitance than a smaller one, with the result
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HANDBOOK
Shunt
+5G +11V
+SG
PARALLEL PLATE FEED
and
Series
Feed
271
+Nv
-BIAS
-BIAS
SERIES PLATE FEED
PARALLEL BIAS FEED
SERIES BIAS FEED
Figure 42
ILLUSTRATING PARALLEL AND
SERIES PLATE FEED
Parallel plate feed is desirable from a safety
standpoint since the tank circuit is at ground
potential with respect to d.c. However, a
Figure 43
ILLUSTRATING SERIES AND
PARALLEL BIAS FEED
high- impedance r-f choke is required, and
the r -t choke must be able to withstand the
peak r -f voltage output of the tube. Series
plate feed eliminates the requirement for a
high -performance r-f choke, but requires the
use of a relatively large value of by-pass
capacitance at the bottom end of the tank
circuit, as contrasted to the moderate value
of coupling capacitance which may be used
at the top of the tank circuit for parallel
plate feed.
that it will actually offer less impedance at
very high frequencies.
Another consideration, just as important as
the amount of d.c. the winding will carry, is
the r -f voltage which may be placed across
the choke without its breaking down. This is
a function of insulation, turn spacing, frequency, number and spacing of pies and other factors.
Some chokes which are designed to have a
high impedance over a very wide range of frequency are, in effect, really two chokes: a
u -h -f choke in series with a high -frequency
choke. A choke of this type is polarized; that
is, it is important that the correct end of the
combination choke be connected to the "hot"
side of the circuit.
Direct-current grid and plate
connections are made either by
series or parallel leed systems.
Simplified forms of each are shown in figures
Shunt and
Series Feed
42 and 43.
Series feed can be defined as that in which
the d -c connection is made to the grid or plate
circuits at a point of very low r-f potential.
Shunt feed always is made to a point of high
r -f voltage and always requires a high impedance r-f choke or a relatively high resistance
to prevent waste of r -f power.
Parallel and
13 -15
Push -Pull Tube Circuits
The comparative r -f power output from parallel or push -pull operated amplifiers is the same
if proper impedance matching is accomplished,
if sufficient grid excitation is available in
both cases, and if the frequency of measurement is considerably lower than the frequency
limit of the tubes.
Operating tubes in parallel has
some advantages in transmitters
designed for operation below 10
NIc., particularly when tetrode or pentode tubes
are to be used. Only one neutralizing capacitor
is required for parallel operation of triode
tubes, as against two for push -pull. Above
about 10 etc., depending upon the tube type,
parallel tube operation is not ordinarily recommended with triode tubes. However, parallel
operation of grounded -grid stages and stages
using low -C beam tetrodes often will give excellent results well into the v -h -f range.
Parallel
Operation
Push -Pull
Operation
The push -pull connection provides
well -balanced circuit insofar as
miscellaneous capacitances are
concerned; in addition, the circuit can be neutralized more completely, especially in high frequency amplifiers. The L/C ratio in a push pull amplifier can be made higher than in a
plate- neutralized parallel -tube operated amplifier. Push -pull amplifiers, when perfectly
balanced, have less second-harmonic output
than parallel or single -tube amplifiers, but in
practice undesired capacitive coupling and
circuit unbalance more or less offset the theoretical harmonic-reducing advantages of push pull r -f circuits.
a
CHAPTER FOURTEEN
R -F
Feedback
Comparatively high gain is required in single sideband equipment because the signal is
usually generated at levels of one watt or less.
To get from this level to a kilowatt requires
about 30 db of gain. High gain tetrodes may
be used to obtain this increase with a minimum
number of stages and circuits. Each stage contributes some distortion; therefore, it is good
practice to keep the number of stages to a
minimum. It is generally considered good practice to operate the low level amplifiers below
their maximum power capability in order to
confine most of the distortion to the last two
amplifier stages. R -f feedback can then be
utilized to reduce the distortion in the last
two stages. This type of feedback is no different from the common audio feedback used
in high fidelity sound systems. A sample of
the output waveform is applied to the amplifier input to correct the distortion developed
in the amplifier. The same advantages can be
obtained at radio frequencies that are obtained
at audio frequencies when feedback is used.
14 -1
R -F Feedback
Circuits
R -f feedback circuits have been developed
by the Collins Radio Co. for use with linear
amplifiers. Tests with large receiving and small
transmitting tubes showed that amplifiers using these tubes without feedback developed
signal -to- distortion ratios no better than 30 db
or so. Tests were run employing cathode follower circuits, such as shown in figure 1A.
Lower distortion was achieved, but at the cost
of low gain per stage. Since the voltage gain
through the tube is less than unity, all gain
has to be achieved by voltage step -up in the
tank circuits. This gain is limited by the dissipation of the tank coils, since the circuit
capacitance across the coils in a typical transmitter is quite high. In addition, the tuning
of such a stage is sharp because of the high
Q circuits.
The cathode follower performance of the
tube can be retained by moving the r -f ground
B
Br
Bi>5
`J
Figure 1
SIMILAR CATHODE FOLLOWER CIRCUITS HAVING DIFFERENT
272
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R -F
GROUND POINTS.
R -F
Feedback Circuits
273
B
B*
BIAS
R -F
Tuning and loading are accomplished by Cr
and C,. C, and L, are tuned in unison to
establish the correct degree of feedback.
R F OUT
BIAS
=
B4
Figure 4
AMPLIFIER WITH FEEDBACK
AND IMPEDANCE MATCHING
OUTPUT NETWORK.
Figure 2
SINGLE STAGE AMPLIFIER WITH
R -F FEEDBACK CIRCUIT
B}
Figure 3
SINGLE STAGE FEEDBACK
AMPLIFIER WITH GROUND
RETURN POINT MODIFIED FOR
UNBALANCED INPUT AND
OUTPUT CONNECTIONS.
point of the circuit from the plate to the cathode as shown in figure 1B. Both ends of the
input circuit are at high r -f potential so inductive coupling to this type of amplifier is
necessary.
Inspection of figure 1B shows that by moving the top end of the input tank down on a
voltage divider tap across the plate tank circuit, the feedback can be reduced from 100%,
as in the case of the cathode follower circuit,
down to any desired value. A typical feedback
circuit is illustrated in figure 2. This circuit
is more practical than those of figure 1, since
the losses in the input tank are greatly reduced.
A feedback level of 12 db may be achieved
as a good compromise between distortion and
stage gain. The voltage developed across C=
will be three times the grid- cathode voltage.
Inductive coupling is required for this circuit, as shown in the illustration.
The circuit of figure 3 eliminates the need
for inductive coupling by moving the r -f
ground to the point common to both tank
circuits. The advantages of direct coupling between stages far outweigh the disadvantages of
having the r -f feedback voltage appear on the
cathode of the amplifier tube.
In order to match the amplifier to a load,
the circuit of figure 4 may be used. The ratio
of XL, to XC, determines the degree of feedback, so it is necessary to tune them in unison
when the frequency of operation is changed.
Tuning and loading functions are accomplished
by varying C2 and G. L5 may also be varied to
adjust the loading.
Feedback Around o
Two -Stage Amplifier
The maximum phase
shift obtainable over
two simple tuned cir-
cuits does not exceed 180 degrees, and feedback around a two stage amplifier is possible.
The basic circuit of a two stage feedback
amplifier is shown in figure 5. This circuit
is a conventional two -stage tetrode amplifier
except that r -f is fed back from the plate
circuit of the PA tube to the cathode of the
driver tube. This will reduce the distortion
1
E
Figure 5
BASIC CIRCUIT OF TWO -STAGE AMPLIFIER WITH R -F FEEDBACK
Feedback voltage is obtained from a voltage divider across the output circuit and
applied directly to the cathode of the first tube. The input tank circuit is thus
outside the feedback loop.
274
R -F
THE RADIO
Feedback
of both tubes as effectively as using individual
feedback loops around each stage, yet will
allow a higher level of overall gain. With
only two tuned circuits in the feedback loop,
it is possible to use 12 to 15 db of feedback
and still leave a wide margin for stability. It
is possible to reduce the distortion by nearly
as many db as are used in feedback. This circuit has two advantages that are lacking in the
single stage feedback amplifier. First, the filament of the output stage can now be operated
at r -f ground potential. Second, any conventional pi output network may be used.
R -f feedback will correct several types of
distortion. It will help correct distortion caused
by poor power supply regulation, too low grid
bias, and limiting on peaks when the plate
voltage swing becomes too high.
Neutralization
The purpose of neutralization of an r -f amplifier
stage is to balance out effects of the grid -plate capacitance coupling in
the amplifier. In a conventional amplifier using a tetrode tube, the effective input capacity
is given by:
ond
R -F
Feedback
Input Capacitance
where: Ci,,
C..
A
=
Cis
+ Cy.
(1
+A
cos e )
= tube input capacitance
= grid -plate capacitance
= voltage amplification from grid
to plate
e
= phase angle of load
In a typical unneutralized tetrode amplifier
having a stage gain of 33, the input capacitance of the tube with the plate circuit in
resonance is increased by 8 µµfd. due to the
unneutralized grid -plate capacitance. This is
unimportant in amplifiers where the gain (A )
remains constant but if the tube gain varies,
serious detuning and r -f phase shift may result.
A grid or screen modulated r -f amplifier is an
example of the case where the stage gain varies from a maximum down to zero. The gain
of a tetrode r -f amplifier operating below plate
current saturation varies with loading so that
if it drives a following stage into grid current
the loading increases and the gain falls off.
The input of the grid circuit is also affected
by the grid -plate capacitance, as shown in this
equation:
Input Resistance
-
27rf
X
C.N (
Asine
)
This resistance is in shunt with the grid
current loading, grid tank circuit losses, and
driving source impedance. When the plate cir-
cuit is inductive there is energy transferred
from the plate to the grid circuit (positive
feedback ) which will introduce negative resistance in the grid circuit. When this shunt
negative resistance across the grid circuit is
lower than the equivalent positive resistance
of the grid loading, circuit losses, and driving
source impedance, the amplifier will oscillate.
When the plate circuit is in resonance
( phase angle equal to zero) the input resistance due to the grid -plate capacitance becomes
infinite. As the plate circuit is tuned to the
capacitive side of resonance, the input resistance becomes positive and power is actually
transferred from the grid to the plate circuit.
This is the reason that the grid current in an
unneutralized tetrode r -f amplifier varies from
a low value with the plate circuit tuned on the
low frequency side of resonance to a high value
on the high frequency side of resonance The
grid current is proportional to the r -f voltage
on the grid which is varying under these conditions. In a tetrode class All amplifier, the
effect of grid -plate feedback can be observed
by placing a r -f voltmeter across the grid circuit and observing the voltage change as the
plate circuit is tuned through resonance.
If the amplifier is over -neutralized, the effects reverse so that with the plate circuit
tuned to the low frequency side of resonance
the grid voltage is high, and on the high frequency side of resonance, it is low.
Amplifier
Neutralization Check
useful "rule of
thumb" method of
checking neutralization of an amplifier stage (assuming that it
is nearly correct to start with) is to tune both
grid and plate circuits to resonance. Then, observing the r -f grid current, tune the plate circuit to the high frequency side of resonance.
If the grid current rises, more neutralization
capacitance is required. Conversely, if the grid
current decreases, less capacitance is needed.
This indication is very sensitive in a neutralized triode amplifier, and correct neutralization exists when the grid current peaks at the
point of plate current dip. In tetrode power
amplifiers this indication is less pronounced.
Sometimes in a supposedly neutralized tetrode
amplifier, there is practically no change in
grid voltage as the plate circuit is tuned
through resonance, and in some amplifiers it
is unchanged on one side of resonance and
drops slightly on the other side. Another observation sometimes made is a small dip in
the center of a broad peak of grid current.
These various effects are probably caused by
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A
HANDBOOK
R -F
R
-F
Figure 7
NEUTRALIZED AMPLIFIER AND
INHERENT FEEDBACK CIRCUIT.
Neutralization
is achieved by varying
the capacity of Cn.
-
coupling from the plate to the grid circuit
through other paths which are not balanced
out by the particular neutralizing circuit used.
Figure 6 shows an r -f amplifier with negative feed of a One -Stage
back. The voltage developed
R -F Amplifier
across G due to the voltage
divider action of G and C,
is introduced in series with the voltage developed across the grid tank circuit and is in
phase- opposition to it. The feedback can be
made any value from zero to 100% by properly choosing the values of C:. and G.
For reasons stated previously, it is necessary
to neutralize this amplifier, and the relationship for neutralization is:
Feedback and
Neutralization
G
==
275
nur
Figure 6
SINGLE STAGE R -F AMPLIFIER
WITH FEEDBACK RATIO OF
DETERMINES
C C to C
C
STAGE NEUTRALIZATION
G
Feedback Circuits
GP
G.
It is often necessary to add capacitance from
plate to grid to satisfy this relationship
Figure 7 is identical to figure 6 except that
it is redrawn to show the feedback inherent in
this neutralization circuit more clearly. G and
C replace G and C., and the main plate tank
tuning capacitance is G. The circuit of figure
7 presents a problem in coupling to the grid
circuit. Inductive coupling is ideal, but the
extra tank circuits complicate the tuning of a
transmitter which uses several cascaded amplifiers with feedback around each one. The
grid could be coupled to a high source impedance such as a tetrode plate, but the driver
then cannot use feedback because this would
cause the source impedance to be low. A possible solution is to move the circuit ground
point from the cathode to the bottom end of
the grid tank circuit. The feedback voltage then
appears between the cathode and ground
( figure 8 ) . The input can be capacitively
coupled, and the plate of the amplifier can
be capacitively coupled to the next stage. Also,
cathode type transmitting tubes are available
that allow the heater to remain at ground po-
tential when r -f is impressed upon the cathode.
The output voltage available with capacity
coupling, of course, is less than the plate cathode r -f voltage developed by the amount
of feedback voltage across G.
14 -2
Feedback and
Neutralization of
Two -Stage
R -F
a
Amplifier
Feedback around two r -f stages has the advantage that more of the rube gain can be
realized and nearly as much distortion reduction can be obtained using 12 db around two
stages as is realized using 12 db around each
of two stages separately. Figure 9 shows a
basic circuit of a two stage feedback amplifier. Inductive output coupling is used, although a pi- network configuration will also
work well. The small feedback voltage required
is obtained from the voltage divider C. - G
and is applied to the cathode of the driver
tube. C. is only a few gcfd., so this feedback
voltage divider may be left fixed for a wide
frequency range. If the combined tube gain is
160, and 12 db of feedback is desired, the ratio
of G to C. is about 40 to 1. This ratio in
practice may be 400 µµtd. to 2.5 µµfd., for
example.
A complication is introduced into this simplified circuit by the cathode -grid capacitance
R
R-F.N
-i
Figure 8
UNBALANCED INPUT AND OUTPUT
CIRCUITS FOR SINGLE -STAGE
R -F AMPLIFIER WITH FEEDBACK
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F
our
276
THE RADIO
Feedback
R -F
Figure 9
TWO -STAGE AMPLIFIER WITH FEEDBACK.
Included
is a
capacitor ,C) for neutralizing the cathode -grid capacity of the first tube.
by capacitor C and V; is neutralized by the correct ratio of C C-.
V.
is
neutralized
,
of the first tube which causes an undersired
coupling to the input grid circuit. It is necessary to neutralize out this capacitance coupling,
as illustrated in figure 9. The relationship for
neutralization is:
G
Cgt
G
G
The input circuit may be made unbalanced
by making C. five times the capacity of G.
This will tend to reduce the voltage across
the coil and to minimize the power dissipated
by the coil. For proper balance in this case,
G must be five times the grid -filament capacitance of the tube.
Except for tubes having extremely small
grid -plate capacitance, it is still necessary to
properly neutralize both tubes. If the ratio of
G to G is chosen to be equal to the ratio of
the grid -plate capacitance to the grid -filament
capacitance in the second tube (Vg), this tube
will be neutralized. Tubes such as a 4X -150A
have very low grid -plate capacitance and probably will not need to be neutralized when used
in the first (V.) stage. If neutralization is
necessary, capacitor G is added for this purpose and the proper value is given by the
following relationship:
CRO
G
_
Cet
G
G
C.
more feedback from the output stage to overcome.
Neutralizing the circuit of
figure 9 balances out coupling between the input
tank circuit and the output tank circuit, but it
does not remove all coupling from the plate
Tests For
Neutralization
circuit to the grid -cathode tube input. This
latter coupling is degenerative, so applying a
signal to the plate circuit will cause a signal
to appear between grid and cathode, even
though the stage is neutralized. A bench test
for neutralization is to apply a signal to the
plate of the tube and detect the presence of a
signal in the grid coil by inductive coupling
to it. No signal will be present when the stage
is neutralized. Of course, a signal could be inductively coupled to the input and neutralization accomplished by adjusting one branch of
the neutralizing circuit bridge (G for example) for minimum signal on the plate circuit.
Neutralizing the cathode -grid capacitance of
the first stage of figure 9 may be accomplished
by applying a signal to the cathode of the tube
and adjusting the bridge balance for minimum
signal on a detector inductively coupled to the
input coil.
Tuning the two -stage
feedback amplifier of
figure 9 is accomplished in an unconventional way because the
output circuit cannot be tuned for maximum
output signal. This is because the output circuit must be tuned so the feedback voltage
applied to the cathode is in -phase with the
input signal applied to the first grid. When
the feedback voltage is not in- phase, the resultTuning
o Two -Stage
Feedback Amplifier
If neither tube requires neutralization, the
bottom end of the interstage tank circuit may
be returned to r -f ground. The screen and
suppressor of the first tube should then be
grounded to keep the tank output capacitance directly across this interstage circuit and
to avoid common coupling between the feedback on the cathode and the interstage circuit.
A slight amount of degeneration occurs in the
first stage since the tube also acts as a grounded
grid amplifier with the screen as the grounded
grid. The p. of the screen is much lower than
that of the control grid so that this effect may
be unnoticed and would only require slightly
ant grid- cathode voltage increases as shown
in figure 10. When the output circuit is
properly tuned, the resultant grid -cathode voltage on the first tube will be at a minimum, and
the voltage on the interstage tuned circuit will
also be at a minimum.
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HANDBOOK
1
VOLTAGE
VOLTAGE - GRID
TO
Neutralization
277
CATHODE
-
INPUT GRID
TO
GROUND
VOLTAGE - CATHODE TO GROUND
(PEEDBACM)
Figure 12
INTERSTAGE CIRCUIT WITH
SEPARATE NEUTRALIZING
AND FEEDBACK CIRCUITS.
(A.
A
B
Figure 10
VECTOR RELATIONSHIP OF
FEEDBACK VOLTAGE
Output Circuit Properly Tuned
Output Circuit Mis -Tuned
The two -stage amplifier may be tuned by
placing a r -f voltmeter across the interstage
tank circuit ( "hot" side to ground) and tuning
the input and interstage circuits for maximum
meter reading, and tuning the output circuit
for minimum meter reading. If the second tube
is driven into the grid current region, the grid
current meter may be used in place of the r -f
voltmeter. On high powered stages where operation is well into the Class AB region, the
plate current dip of the output tube indicates
correct output circuit tuning, as in the usual
amplifier.
Quite often low freq u e n c y parasitics
may be found in
the interstage circuit of the two -stage feedback
amplifier. Oscillation occurs in the first stage
due to low frequency feedback in the cathode
circuit. R -f chokes, coupling capacitors, and
bypass capacitors provide the low frequency
tank circuits. When the feedback and second
stage neutralizing circuits are combined, it is
necessary to use the configuration of figure 11.
This circuit has the advantage that only one
capacitor (G) is required from the plate of
the output tube, thus keeping the added capacitance across the output tank at a minimum.
Parasitic Oscillations in
the Feedback Amplifier
It is convenient, however, to separate these circuits so neutralization and feedback can be
adjusted independently. Also, it may be desirable to be able to switch the feedback out
of the circuit. For these reasons, the circuit
shown in figure 12 is often used. Switch S1
removes the feedback loop when it is closed.
A slight tendency for low frequency parasitic oscillations still exists with this circuit.
Li should have as little inductance as possible
without upsetting the feedback. If the value of
Li is too low, it cancels out part of the reactance of feedback capacitor G and causes
the feedback to increase at low values of radio
frequency. In some cases, a swamping resistor
may be necessary across L. The value of this
resistor should be high compared to the reactance of G to avoid phase -shift of the r -f
feedback.
Neutralization
14 -3
Procedure in
Feedback -Type Amplifiers
Experience with feedback amplifiers has
brought out several different methods of neutralizing. An important observation is that
when all three neutralizing adjustments are
correctly made the peaks and dips of various
tuning meters all coincide at the point of circuit resonance. For example, the coincident indications when the various tank circuits are
tuned through resonance with feedback operating are:
A -When the PA plate circuit is tuned
through resonance:
-PA plate current dip
2 -Power output peak
3 -PA r -f grid voltage dip
4 -PA grid current dip
1
+
BAS
Figure 11
INTERSTAGE CIRCUIT COMBINING
NEUTRALIZATION AND
FEEDBACK NETWORKS.
(Note: The PA grid current peaks
when feedback circuit is disabled
and the tube is heavily driven)
278
R -F
R-F
THE RADIO
Feedback
INc
Li o
O
°
cll
p FOUT
11(-
pi.
TY
gifIIN 1 c
C:7T
F
Tc.
ÌIr
ct0
HI,
your
RFC
1- 1-
o
T
1
l-
6+
BIAS
BIAS
Figure 13
TWO -STAGE AMPLIFIER WITH
B-When the PA grid circuit is tuned
through resonance:
Driver plate current dip
2 -PA r -f grid voltage peak
3 -PA grid current peak
4 -PA power output peak
1-
C -When the driver grid circuit is tuned
through resonance:
T
°
GF
FEED BACK
CIRCUIT.
2- Neutralize the grid -plate
tance of the driver stage
3- Neutralize the grid -plate
capaci-
capacitance of the power amplifier (PA)
stage
4 -Apply r -f feedback
Neutralize driver grid- cathode capacitance
5-
1-Driver r -f grid voltage peak
2-Driver plate current peak
These steps will be explained in more detail
in the following paragraphs:
3
Step 1. The removal of r -f feedback through
the feedback circuit must be complete. The
switch ( ) shown in the feedback circuit
( figure 13 ) is one satisfactory method. Since
C. is effectively across the PA plate tank circuit it is desirable to keep it across the circuit
when feedback is removed to avoid appreciable
detuning of the plate tank circuit. Another
method that can be used if properly done is
to ground the junction of C and C. Grounding this common point through a switch or
relay is not good enough because of common
coupling through the length of the grounding
lead. The grounding method shown in figure
14 is satisfactory.
-PA r -f grid current peak
-PA plate current peak
5 -PA power output peak
4
Four meters may be employed to measure
the most important of these parameters. The
meters should be arranged so that the following pairs of readings are displayed on meters
located close together for ease of observation
of coincident peaks and dips:
2
-PA plate current and power output
-PA r -f grid current and PA plate
3
-PA r -f grid
1
current
voltage and power output
Driver plate current and PA r-f
grid voltage
4-
The third pair listed above may not be
necessary if the PA plate current dip is pronounced. When this instrumentation is provided, the neutralizing procedure is as follows:
1- Remove the r-f feedback
FEEDBACK
Figure 14
SHORTING DEVICE.
Step 2. Plate power and excitation are applied.
The driver grid tank is resonated by tuning
for a peak in driver r -f grid voltage or driver
plate current. The power amplifier grid tank
circuit is then resonated and adjusted for a
dip in driver plate current. Driver neutralization is now adjusted until the PA r -f grid
voltage (or PA grid current) peaks at exactly
the point of driver plate current dip. A handy
rule for adjusting grid-plate neutralization of
a tube without feedback: with all circuits in
resonance, detune the plate circuit to the high
frequency side of resonance: If grid current
to next stage (or power output of the stage
under test) increases, more neutralizing capacitance is required and vice versa.
If the driver tube operates class A so that
a plate current dip cannot be observed, a dif-
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HANDBOOK
Neutralization
279
Neutralization
The method of neutralization
employing a sensitive r -f detector inductively coupled to
a tank coil is difficult to apply in some cases
because of mechanical construction of the
equipment, or because of undesired coupling.
Another method for observing neutralization
can be used, which appears to be more accurate in actual practice. A sensitive r -f detector such as a receiver is loosely coupled to the
grid of the stage being neutralized, as shown
in figure 15. The coupling capacitance is of
the order of one or two µµfd. It must be small
enough to avoid upsetting the neutralization
when it is removed because the total grid ground capacitance is one leg of the neutralizing bridge. A signal generator is connected at
point S and the receiver at point R. If Coo is
not properly adjusted the S -meter on the receiver will either kick up or down as the grid
tank circuit is tuned through resonance. Go
may be adjusted for minimum deflection of the
S-meter as the grid circuit is tuned through
Techniques
T"
Figure 15
FEEDBACK NEUTRALIZING
CIRCUIT USING
AUXILIARY RECEIVER.
ferent neutralizing procedure is necessary This
will be discussed in a subsequent section.
Step 3. This is the same as step 2 except it
is applied to the power amplifier stage. Adjust the neutralization of this stage for a peak
in power output at the plate current dip.
Step 4. Reverse step
back.
1
and apply the r -f feed-
resonance.
Step 5. Apply plate power and an exciting signal to drive the amplifier to nearly full output. Adjust the feedback neutralization for a
peak in amplifier power output at the exact
point of minimum amplifier plate current.
Decrease the feedback neutralization capacitance if the power output rises when the tank
circuit is tuned to the high frequency side of
resonance.
The above sequence applies when the neutralizing adjustments are approximately correct to start with. If they are far off, some "cut and -try" adjustment may be necessary. Also,
the driver stage may break into oscillation if
the feedback neutralizing capacitance is not
near the correct setting.
It is assumed that a single tone test signal
for amplifier excitation during the
above steps, and that all tank circuits are at
resonance except the one being detuned to
make the observation. There is some interaction
between the driver neutralization and the feedback neutralization so if an appreciable change
is made in any adjustment the others should
be rechecked. It is important that the grid -plate
neutralization be accomplished first when using
is used
the above procedure, otherwise the feedback
neutralization will be off a little, since it partially compensates for that error.
The grid-plate capacitance of the tube is
then neutralized by connecting the signal generator to the plate of the tube and adjusting
Cti of figure 13 for minimum deflection again
as the grid tank is tuned through resonance.
The power amplifier stage is neutralized in
the same manner by connecting a receiver
loosely to the grid circuit, and attaching a
signal generator to the plate of the tube. The
r -f signal can be fed into the amplifier output
terminal if desired.
Some precautions are necessary when using
this neutralization method. First, some driver
tubes (the 6CL6, for example) have appreciably more effective input capacitance when
in operation and conducting plate current than
when in standby condition. This increase in
input capacitance may be as great as three or
four µµfd, and since this is part of the neutralizing bridge circuit it must be taken into
consideration. The result of this change in
input capacitance is that the neutralizing adjustment of such tubes must be made when
they are conducting normal plate current. Stray
coupling must be avoided, and it may prove
helpful to remove filament power from the
preceding stage or disable its input circuit in
some manner.
It should be noted that in each of the above
adjustments that minimum reaction on the
grid is desired, not minimum voltage. Some
residual voltage is inherent on the grid when
this neutralizing circuit is used.
CHAPTER FIFTEEN
Amplitude Modulation
If the output of a c -w transmitter is varied
in amplitude at an audio frequency rate instead of interrupted in accordance with code
characters, a tone will be heard on a receiver
tuned to the signal. If the audio signal consists of a band of audio frequencies comprising voice or music intelligence, then the
voice or music which is superimposed on the
radio frequency carrier will be heard on the
receiver.
When voice, music, video, or other intelligence is superimposed on a radio frequency
carrier by means of a corresponding variation
in the amplitude of the radio frequency output
of a transmitter, amplitude modulation is the
result. Telegraph keying of a c -w transmitter
is the simplest form of amplitude modulation,
while video modulation in a television transmitter represents a highly complex form. Systems for modulating the amplitude of a carrier
envelope in accordance with voice, music, or
similar types of complicated audio waveforms
are many and varied, and will be discussed
later on in this chapter.
15-1
Sidebands
Modulation is essentially a form of mixing
or combining already covered in a previous
chapter. To transmit voice at radio frequencies
by means of amplitude modulation, the voice
frequencies are mixed with a radio frequency
carrier so that the voice frequencies are converted to radio frequency sidebands. Though
it may be difficult to visualize, the amplitude
of the radio frequency carrier does not vary
during conventional amplitude modulation.
Even though the amplitude of radio frequency voltage representing the composite
signal ( resultant of the carrier and sidebands,
called the envelope) will vary from zero to
twice the unmodulated signal value during
full modulation, the amplitude of the carrier
component does not vary. Also, so long as
the amplitude of the modulating voltage does
not vary, the amplitude of the sidebands will
remain constant. For this to be apparent, however, it is necessary to measure the amplitude
of each component with a highly selective
filter. Otherwise, the measured power or voltage will be a resultant of two or more of the
components, and the amplitude of the resultant
will vary at the modulation rate.
If a carrier frequency of 5000 kc. is modulated by a pure tone of 1000 cycles, or 1 kc.,
two sidebands are formed: one at 5001 kc.
(the sum frequency) and one at 4999 kc. (the
difference frequency). The frequency of each
sideband is independent of the amplitude of
the modulating tone, or modulation percentage; the frequency of each sideband is determined only by the frequency of the modulating tone. This assumes, of course, that the
transmitter is not modulated in excess of its
linear capability.
280
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Modulation
When the modulating signal consists of
multiple frequencies, as is the case with
voice or music modulation, two sidebands will
be formed by each modulating frequency (one
on each side of the carrier), and the radiated
signal will consist of a band of frequencies.
The band width, or channel taken up in the
frequency spectrum by a conventional double sideband amplitude-modulated signal, is equal
to twice the highest modulating frequency.
For example, if the highest modulating frequency is 5000 cycles, then the signal (assuming modulation of complex and varying
waveform) will occupy a band extending from
5000 cycles below the carrier to 5000 cycles
above the carrier.
Frequencies up to at least 2500 cycles, and
preferably 3500 cycles, are necessary for good
speech intelligibility. If a filter is incorporated in the audio system to cut out all frequencies above approximately 3000 cycles,
the band width of a radio- telephone signal can
be limited to 6 kc. without a significant loss
in intelligibility. However, if harmonic distortion is introduced subsequent to the filter, as
would happen in the case of an overloaded
modulator or overmodulation of the carrier,
new frequencies will be generated and the
signal will occupy a band wider than 6 kc.
Mechanics of
Modulation
15-2
fl
281
f
A
C.W. OR UNMODULATED CARRIER
SINE WAVE
AUDIO SIGNAL FROM MODULATOR
A 2
ItiÌ% Ì 1TjTI1 1ZIUIÌ
1111111111111111111111111
lA/2
_A /2
III 111
ÌI,
50
t
% MODULATED CARRIER
A
A
A
00%
MODULATED CARRIER
Figure
AMPLITUDE MODULATED WAVE
Top drawing (Al represents an unmodulated
carrier wave; (B) shows the audio output of
the modulator. Drawing (C) shows the audio
signal impressed on the carrier wave to the
extent of 50 per cent modulation; (D) shows
the carrier with 100 per cent amplitude modulation.
1
A c -w or unmodulated r-f carrier wave is
represented in figure 1A. An audio frequency
sine wave is represented by the curve of
figure
113.
When
the two are combined or
"mixed," the carrier is said to be amplitude
modulated, and a resultant similar to 1C or
is obtained. It should be noted that under
modulation, each half cycle of r -f voltage
differs slightly from the preceding one and
the following one; therefore at no time during
modulation is the r-f waveform a pure sine
wave. This is simply another way of saying
that during modulation, the transmitted r -f
energy no longer is confined to a single radio
frequency.
It will be noted that the average amplitude
of the peak r -f voltage, or modulation envelope, is the same with or without modulation.
This simply means that the modulation is
symmetrical (assuming a symmetrical modulating wave) and that for distortionless modulation the upward modulation is limited to a
value of twice the unmodulated carrier wave
amplitude because the amplitude cannot go
below zero on downward portions of the modulation cycle. Figure 1D illustrates the maxi1D
obtainable distortionless modulation with
a sine modulating wave, the r -f voltage at the
peak of the r -f cycle varying from zero to
twice the unmodulated value, and the r -f power
varying from zero to four times the unmodulated value ( the power varies as the square
of the voltage).
While the average r -f voltage of the modulated wave over a modulation cycle is the
mum
same as for the unmodulated carrier, the average power increases with modulation. If the
radio frequency power is integrated over the
audio cycle, it will be found with 100 per cent
sine wave modulation the average r-f power
has increased 50 per cent. This additional
power is represented by the sidebands, because as previously mentioned, the carrier
power does not vary under modulation. Thus,
when a 100 -watt carrier is modulated 100 per
cent by a sine wave, the total r -f power is 150
watts; 100 watts in the carrier and 25 watts
in each of the two sidebands.
www.americanradiohistory.com
282
Amplitude Modulation
THE
long as the relative proporLion of the various sidebands
Modulation
Percentage
RADIO
So
making up voice modulation is
maintained, the signal may be received and
detected without distortion. However, the
higher the average amplitude of the sidebands,
the greater the audio signal produced at the
receiver. For this reason it is desirable to
increase the modulation percentage, or degree
of modulation, to the point where maximum
peaks just hit 100 per cent. If the modulation
percentage is increased so that the peaks exceed this value, distortion is introduced, and
if carried very far, bad interference to signals
on nearby channels will result.
The amount by which a carrier
is being modulated may be expressed either as a modulation
factor, varying from zero to 1.0 at maximum
modulation, or as a percentage. The percentage of modulation is equal to 100 times the
modulation factor. Figure 2A shows a carrier
wave modulated by a sine -wave audio tone.
A picture such as this might be seen on the
screen of a cathode -ray oscilloscope with
sawtooth sweep on the horizontal plates and
the modulated carrier impressed on the vertical plates. The same carrier without modulation
would appear on the oscilloscope screen as
ECAR
Figure
2
GRAPHICAL DETERMINATION OF MODULATION PERCENTAGE
The procedure for determining modulation
percentage from the peak voltage points indicated is discussed in the text.
Modulation
Measurement
figure 2B.
The percentage of modulation of the positive peaks and the percentage of modulation
of the negative peaks can be determined separately from two oscilloscope pictures such
as shown.
The modulation factor of the positive peaks
may be determined by the formula:
Emax
M
=
-
Ecar
Ecar
The factor for negative peaks may be determined from this formula:
M
-
Ecar
-
Emin
Ecar
In the above two formulas Ern ax is the maximum carrier amplitude with modulation and
Ellin is the minimum amplitude; Ecar is the
steady -state amplitude of the carrier without modulation. Since the deflection of the
spot on a cathode -ray tube is linear with respect to voltage, the relative voltages of
these various amplitudes may be determined
by measuring the deflections, as viewed on
the screen, with a rule calibrated in inches
or centimeters. The percentage of modulation
of the carrier may be had by multiplying the
modulation factor thus obtained by 100. The
above procedure assumes that there is no
carrier shift, or change in average amplitude,
with modulation.
If the modulating voltage is symmetrical,
such as a sine wave, and modulation is accomplished without the introduction of distortion, then the percentage modulation will
be the same for both negative and positive
peaks. However, the distribution and phase
relationships of harmonics in voice and music
waveforms are such that the percentage modulation of the negative modulation peaks may
exceed the percentage modulation of the positive peaks, and vice versa. The percentage
modulation when referred to without regard
to polarity is an indication of the average of
the negative and positive peaks.
The modulation capability of a
transmitter is the maximum percentage to which that transmitter
may be modulated before spurious sidebands
are generated in the output or before the distortion of the modulating waveform becomes
objectionable. The highest modulation capability which any transmitter may have on the
negative peaks is 100 per cent. The maximum
permissible modulation of many transmitters
is less than 100 per cent, especially on positive peaks. The modulation capability of a
transmitter may be limited by tubes with inModulation
Capability
sufficient filament emission, by insufficient
excitation or grid bias to a plate- modulated
stage, too light loading of any type of amplifier carrying modulated r.f., insufficient power
output capability in the modulator, or too much
excitation to a grid-modulated stage or a
Class B linear amplifier. In any case, the
FCC regulations specify that no transmitter
be modulated in excess of its modulation
capability. Hence, it is desirable to make the
modulation capability of a transmitter as near
as possible to 100 per cent so that the carrier
power may be used most effectively.
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HANDBOOK
Speech Waveform
Modulation Systems
The manner in which the
human voice is produced
by the vocal cords gives
rise to a certain dissymmetry in the waveform
of voice sounds when they are picked up by
a good -quality microphone. This is especially
pronounced in the male voice, and more so
on certain voiced sounds than on others. The
result of this dissymmetry in the waveform is
that the voltage peaks on one side of the
average value of the wave will be considerably greater, often two or three times as great,
as the voltage excursions on the other side
of the zero axis. The average value of voltage on both sides of the wave is, of course,
the same.
As a result of this dissymmetry in the male
voice waveform, there is an optimum polarity
of the modulating voltage that must be observed if maximum sideband energy is to be
obtained without negative peak clipping and
generation of "splatter" on adjacent channels.
A double -pole double -throw "phase reversing" switch in the input or output leads of any
transformer in the speech amplifier system will
permit poling the extended peaks in the direction of maximum modulation capability. The
optimum polarity may be determined easily by
listening on a selective receiver tuned to a
frequency 30 to 50 kc. removed from the desired signal and adjusting the phase reversing
switch to the position which gives the least
"splatter" when the transmitter is modulated
rather heavily. If desired, the switch then may
be replaced with permanent wiring, so long as
the microphone and speech system are not to
be changed.
A more conclusive illustration of the lopsidedness of a speech waveform may be obtained by observing the modulated waveform
of a radiotelephone transmitter on an oscilloscope. A portion of the carrier energy of the
transmitter should be coupled by means of a
link directly to the vertical plates of the
'scope, and the horizontal sweep should be a
sawtooth or similar wave occurring at a rate
of approximately 30 to 70 sweeps per second.
With the speech signal from the speech amplifier connected to the transmitter in one polarity it will be noticed that negative -peak
clipping -as indicated by bright "spots" in
the center of the 'scope pattern whenever the
carrier amplitude goes to zero -will occur at
a considerably lower level of average modulation than with the speech signal being fed to
the transmitter in the other polarity. When the
input signal to the transmitter is polarized in
such a manner that the "fingers" of the
speech wave extend in the direction of positive modulation these fingers usually will be
clipped in the plate circuit of the modulator
at an acceptable peak modulation level.
Dissymmetry
283
The use of the proper polarity of the incoming speech wave in modulating a transmitter
can afford an increase of approximately two
to one in the amount of speech audio power
which may be placed upon the carrier for an
amplitude-modulated transmitter for the same
amount of sideband splatter. More effective
methods for increasing the amount of audio
power on the carrier of an AM phone transmitter are discussed later in this chapter.
Because the same intelligibility is contained in each
of the sidebands associated
with a modulated carrier, it is not necessary
to transmit sidebands on both sides of the
carrier. Also, because the carrier is simply a
single radio frequency wave of unvarying amplitude, i t is no t necessary to transmit the
carrier if some means is provided for inserting
a locally generated carrier at the receiver.
When the carrier is suppressed but both
upper and lower sidebands are transmitted, it
is necessary to insert a locally generated
carrier at the receiver of exactly the same
frequency and phase as the carrier which was
suppressed. For this reason, suppressed carrier double -sideband systems have little
practical application.
When the carrier is suppressed and only the
upper or the lower sideband is transmitted, a
highly intelligible signal may be obtained at
the receiver even though the locally generated
carrier differs a few cycles from the frequency
of the carrier which was suppressed at the
transmitter. A communications system utilizing but one group of sidebands with carrier
suppressed is known as a single sideband
system. Such systems are widely used for
commercial point to point work, and are being
used to an increasing extent in amateur communication. The two chief advantages of the
system are: (1) an effective power gain of
about 9 db results from putting all the radiated power in intelligence carrying sideband
frequencies instead of mostly into radiated
carrier, and (2) elimination of the selective
fading and distortion that normally occurs in
a conventional double - sideband system when
the carrier fades and the sidebands do not, or
the sidebands fade differently.
Single- Sideband
Transmission
15 -3
Systems of Amplitude
Modulation
There are many different systems and methods for amplitude modulating a carrier, but
most may be grouped under three general classifications: (1) variable efficiency systems
in which the average input to the stage re-
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284
THE
Amplitude Modulation
mains constant with and without modulation
and the variations in the efficiency of the
stage in accordance with the modulating signal accomplish the modulation; (2) constant
efficiency systems in which the input to the
stage is varied by an external source of modulating energy to accomplish the modulation;
and (3) so- called high -efficiency systems in
which circuit complexity is increased to obhigh plate circuit efficiency in the modulated
stage without the requirement of an external
high -level modulator. The various systems
under each classification have individual
characteristics which make certain ones best
suited to particular applications.
Since the average input
remains constant in a
stage employing variable
efficiency modulation, and since the average
power output of the stage increases with modulation, the additional average power output
from the stage with modulation must come from
the plate dissipation of the tubes in the stage.
Hence, for the best relation between tube cost
and power output the tubes employed should
have as high a plate dissipation rating per
Variable Efficiency
Modulation
dollar as possible.
The plate efficiency in such
an
ciency in certain types of amplifiers will be
as low as 60 per cent, the unmodulated efficiency in such amplifiers will be in the vicinity of 30 per cent.
Assuming a typical amplifier having a peak
efficiency of 70 per cent, the following figures give an idea of the operation of an idealized efficiency -modulated stage adjusted for
100 per cent sine -wave modulation. It should
be kept in mind that the plate voltage is constant at all times, even over the audio cycles.
100 watts
35 watts
35%
Input on 100% positive modulation
peak (plate current doubles)
Efficiency on
200 watts
70%
Output
tion peak
140
watts
0
watts
100% positive peak
on 100% positive modula-
Input on 100% negative peak
Efficiency on 100% negative peak
Output on 100% negative peak
0%
0 watts
100
watts
52.5 watts
52.5%
Systems of Efficiency
There are many systems of efficiency modulation, but they all
have the general limitation discussed in the
previous paragraph -so long as the carrier
amplitude is to remain constant with and
without modulation, the efficiency at carrier
level must be not greater than one -half the
peak modulation efficiency if the stage is to
be capable of 100 per cent modulation.
The classic example of efficiency modulation is the Class B linear r -f amplifier, to be
discussed below. The other three common
forms of efficiency modulation are control grid modulation, screen -grid modulation, and
suppressor -grid modulation. In each case,
including that of the Class B linear amplifier
note that the modulation, or the modulates
signal, is impressed on a control electrode
of the stage.
Modulation
amplifier is
when going from the unmodulated
condition to the peak of the modulation cycle.
Hence, the unmodulated efficiency of such an
amplifier must always be less than 45 per
cent, since the maximum peak efficiency obtainable in a conventional amplifier is in the
vicinity of 90 per cent. Since the peak effi-
doubled
Plate input without modulation
Output without modulation
Efficiency without modulation
Average input with 100%
modulation
Average output with 100% modulation (35 watts carrier plus 17.5
watts sideband)
Average efficiency with 100%
modulation
RADIO
The Class B
Linear Amplifier
This is the simplest practicable type amplifier for an
amplitude -modulated wave
or a single -sideband signal. The system possesses the disadvantage that excitation, grid
bias, and loading must be carefully controlled
to preserve the linearity of the stage. Also,
the grid circuit of the tube, in the usual application where grid current is drawn on peaks,
presents a widely varying value of load impedance to the source of excitation. Hence it
is necessary to include some sort of swamping
resistor to reduce the effect of grid- impedance variations with modulation. If such a
swamping resistance across the grid tank is
not included, or is too high in value, the positive modulation peaks of the incoming modulated signal will tend to be flattened with
resultant distortion of the wave being amplified.
The Class B linear amplifier has long been
used in broadcast transmitters, but recently
has received much more general usage in the
h -f range for two significant reasons: (a) the
Class B linear is an excellent way of increasing the power output of a single -sideband
transmitter, since the plate efficiency with
full signal will be in the vicinity of 70 per
cent, while with no modulation the input to
the stage drops to a relatively low value; and
(b) the Class B linear amplifier operates with
relatively low harmonic output since the grid
bias on the stage normally is slightly less
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HANDBOOK
Class
than the value which will cut off plate current
to the stage in the absence of excitation.
Since s Class B linear amplifier is biased
to extended cutoff with no excitation ( the
grid bias at extended cutoff will be approximately equal to the plate voltage divided by
the amplification factor for a triode, and will
be approximately equal to the screen voltage
divided by the grid- screen mu factor for a
tetrode or pentode) the plate current will flow
essentially in 180 -degree pulses. Due to the
relatively large operating angle of plate current flow the theoretical peak plate efficiency
is limited to 78.5 per cent, with 65 to 70 per
cent representing a range of efficiency normally attainable, and the harmonic output
peak output of the r -f envelope should fall to
half the value obtained on positive modula-
will
be low.
The carrier power output from a Class B
linear amplifier of a normal 100 per cent modulated AM signal will be about one -half the
rated plate dissipation of the stage, with optimum operating conditions. The peak output
from a Class B linear, which represents the
maximum- signal output as a single -sideband
amplifier, or peak output with a 100 per cent
AM signal, will be about twice the plate dissipation of the tubes in the stage. Thus the
carrier -level input power to a Class B linear
should be about 1.5 times the rated plate dissipation of the stage.
The schematic circuit of a Class B linear
amplifier is the same as a conventional singleended or push -pull stage, whether triodes or
beam tetrodes are used. However, a swamping
resistor, as mentioned before, must be placed
across the grid tank of the stage if the operating conditions of the tube are such that
appreciable gridcurrent will be drawn on modulation peaks. Also, a fixed source of grid bias
must be provided for the stage. A regulated
grid -bias power supply is the usual source of
negative bias voltage.
With grid bias adjusted
to the correct value,
and with provision for
varying the excitation voltage to the stage
and the loading of the plate circuit, a fully
modulated signal is applied to the grid circuit
of the stage. Then with an oscilloscope coupled to the output of the stage, excitation and
loading are varied until the stage is drawing
the normal plate input and the output wave shape is a good replica of the input signal.
The adjustment procedure normally will reAdjustment of a Class
8 Linear Amplifier
quire a succession of approximations, until
the optimum set of adjustments is attained.
Then the modulation being applied to the input signal should be removed to check the
linearity. With modulation removed, in the
case of a 100 per cent AM signal, the input
to the stage should remain constant, and the
B
Linear Amplifier
285
tion peaks.
Class C
Grid Modulation
.
One widely used system of
efficiency
modulation for
communications
work
is
Class C control -grid bias modulation. The distortion is slightly higher than for a properly
operated Class B linear amplifier, but the efficiency is also higher, and the distortion can
be kept within tolerable limits for communications work.
Class C grid modulation requires high plate
voltage on the modulated stage, if maximum
output is desired. The plate voltage is normally run about 50 per cent higher than for
maximum output with plate modulation.
The driving power required for operation of
a grid -modulated amplifier under these conditions is somewhat more than is required for
operation at lower bias and plate voltage, but
the increased power output obtainable overbalances the additional excitation requirement. Actually, almost half as much excitation
is required as would be needed if the same
stage were to be operated as a Class C plate modulated amplifier. The resistor R across
the grid tank of the stage serves as swamping
to stabilize the r -f driving voltage. At least
50 per cent of the output of the driving stage
should be dissipated in this swamping resistor
under carrier conditions.
A comparatively small amount of audio power
will be required to modulate the amplifier stage
100 per cent. An audio amplifier having 20
watts output will be sufficient to modulate an
amplifier with one kilowatt input. Proportionately smaller amounts of audio will be required for lower powered stages. However, the
audio amplifier that is being used as the grid
modulator should, in any case, either employ
low plate resistance tubes such as 2A3's,
employ degenerative feedback from the output
stage to one of the preceding stages of the
speech amplifier, or be resistance loaded with
a resistor across the secondary of the modulation transformer. This provision of low drive
ing impedance in the grid modulator is to insure
good regulation in the audio driver for the grid
modulated stage. Good regulation of both the
audio and the r -f drivers of a grid -modulated
stage is quite important if distortion-free
modulation approaching 100 per cent is desired,
because the grid impedance of the modulated
stage varies widely over the audio cycle.
A practical circuit for obtaining grid -bias
modulation is shown in figure 3. The modulator and bias regulator tube have been combined in a single 6B4G tube.
The regulator -modulator tube operates as
a cathode - follower. The average d -c voltage
www.americanradiohistory.com
THE
Amplitude Modulation
286
R.F. AMPLIFIER
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RFC
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The most satisfactory pro cedure for tuning a stage
for grid -bias modulation of
the Class C type is as
follows. The amplifier should first be neutralized, and any possible tendency toward parasitics under any condition of operation should
be eliminated. Then the antenna should be
coupled to the plate circuit, the grid bias
should be run up to the maximum available
value, and the plate voltage and excitation
should be applied. The grid bias voltage
should then be reduced until the amplifier
draws the approximate amount of plate current it is desired to run, and modulation corresponding to about 80 per cent is then applied.
If the plate current kicks up when modulation
is applied, the grid bias should be reduced;
if the plate meter kicks down, increase the
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TRANSFORMER
3
GRID -BIAS MODULATOR
per cent. If the antenna coupling is decreased
slightly from the condition just described, and
the excitation is increased to the point where
the amplifier draws the same input, carrier
efficiency of 50 per cent is obtainable with
tolerable distortion at 90 per cent modulation.
Tuning the
Grid -Bias
Modulated Stage
*5
25K IOW
AUDIO INPUT
RADIO
CIRCUIT
on the control grid is controlled by the 70, 000 ohm wire -wound potentiometer and this potentiometer adjusts the average grid bias on the
modulated stage. However, a -c signal voltage
is also impressed on the control -grid of the
tube and since the cathode follows this a -c
wave the incoming speech wave is superimposed on the average grid bias, thus effecting
grid -bias modulation of the r -f amplifier stage.
An audio voltage swing is required on the grid
of the 6B4G of approximately the same peak
value as will be required as bias -voltage
swing on the grid -bias modulated stage. This
voltage swing will normally be in the region
from 50 to 200 peak volts. Up to about 100
volts peak swing can be obtained from a 6SJ7
tube as a conventional speech amplifier stage.
The higher voltages may be obtained from a
tube such as a 6J5 through an audio transformer of 2:1 or 21/3:1 ratio.
With the normal amount of comparatively
tight antenna coupling to the modulated stage,
a non -modulated carrier efficiency of 40 per
cent can be obtained with substantially distortion -free modulation up to practically 100
grid bias.
When the amount of bias voltage has been
found (by adjusting the fine control, R2, on
the bias supply) where the plate meter remains constant with modulation, it is more
than probable that the stage will be drawing
either too much or too little input. The antenna coupling should then be either increased
or decreased (depending on whether the input was too little or too much, respectively)
until the input is more nearly the correct value.
The bias should then be readjusted until the
plate meter remains constant with modulation
as before. By slight jockeying back and forth
of antenna coupling and grid bias, a point can
be reached where the tubes are running at
rated plate dissipation, and where the plate
milliammeter on the modulated stage remains
substantially constant with modulation.
The linearity of the stage should then
be
checked by any of the conventional methods;
the trapezoidal pattern method employing a
cathode -ray oscilloscope is probably the most
satisfactory. The check with the trapezoidal
pattern will allow the determination of the
proper amount of gain to employ on the speech
amplifier. Too much audio power on the grid
of the modulated stage should not be used in
the tuning -up process, as the plate meter will
kick erratically and it will be impossible to
make a satisfactory adjustment.
Amplitude modulation may be
accomplished by varying the
screen -grid voltage in a Class
amplifier which employs a pentode, beam
Screen -Grid
Modulation
C
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Screen Grid Modulation
H A N D B O O K
tetrode, or other type of screen -grid tube. The
modulation obtained in this way is not especially linear, but screen -grid modulation
does offer other advantages and the linearity
is quite adequate for communications work.
There are two significant and worthwhile
advantages of screen -grid modulation for communications work: (1) The excitation requirements for an amplifier which is to be modulated in the screen are not at all critical, and
good regulation of the excitation voltage is
not required. The normal rated grid- circuit
operating conditions specified for Class C
c -w operation are quite adequate for screen grid modulation. (2) The audio modulating
power requirements for screen -grid modulation
are relatively low.
A screen -grid modulated r -f amplifier operates as an efficiency -modulated amplifier, the
same as does a Class B linear amplifier and
a grid -modulated stage. Hence, plate circuit
loading is relatively critical as in any efficiency- modulated stage, and must be adjusted
to the correct value if normal power output
with full modulation capability is to be obtained. As in the case of any efficiency -modulated stage, the operating efficiency at the
peak of the modulation cycle will be between
70 and 80 per cent, with efficiency at the carrier level (if the stage is operating in the normal manner with full carrier) about half of the
peak- modulation value.
There are two main disadvantages of screen grid modulation, and several factors which
must be considered if satisfactory operation
of the screen -grid modulated stage is to be
obtained. The disadvantages are: (I) As mentioned before, the linearity of modulation with
respect to screen -grid voltage of such a stage
is satisfactory only for communications work,
unless carrier- rectified degenerative feed -back
is employed around the modulated stage to
straighten the linearity of modulation. (2) The
impedance of the screen grid to the modulating
signal is non -linear. This means that the modulating signal must be obtained from a source
of quite low impedance if audio distortion of
the signal appearing at the screen grid is to
be avoided.
Instead of being linear with respect to modulating voltage, as
is the plate circuit of a plate modulated Class C amplifier, the screen grid
presents approximately a square-law impedance to the modulating signal over the region
of signal excursion where the screen is positive with respect to ground. This non -linearity
may be explained in the following manner: At
the carrier level of a conventional screen modulated stage the plate -voltage swing of
the modulated tube is one -half the voltage
Screen -Grid
Impedance
287
swing at peak- modulation level. This condition
must exist in any type of conventional efficiency- modulated stage if 100 per cent positive modulation is to be attainable. Since the
plate -voltage swing is at half amplitude, and
since the screen voltage is at half its full modulation value, the screen current is relatively low. But at the positive modulation peak
the screen voltage is approximately doubled,
and the plate -voltage swing also is at twice
the carrier amplitude. Due to the increase in
plate -voltage swing with increasing screen
voltage, the screen current increases more than
linearly with increasing screen voltage.
In a test made on an amplifier with an 813
tube, the screen current at carrier level was
about 6 ma. with screen potential of 190 volts;
but under conditions which represented a positive modulation peak the screen current measured 25 ma. at a potential of 400 volts. Thus
instead of screen current doubling with twice
screen voltage as would be the case if the
screen presented a resistive impedance, the
screen current became about four times as
great with twice the screen voltage.
Another factor which must be considered
in the design of a screen -modulated stage, if
full modulation is to be obtained, is that the
power output of a screen -grid stage with zero
screen voltage is still relatively large. Hence,
if anything approaching full modulation on
negative peaks is to be obtained, the screen
potential must be made negative with respect
to ground on negative modulation peaks. In
the usual types of beam tetrode tubes the
screen potential must be 20 to 50 volts negative with respect to ground before cut -off of
output is obtained. This condition further complicates the problem of obtaining good linearity
in the audio modulating voltage for the screen modulated stage, since the screen voltage
must be driven negatively with respect to
ground over a portion of the cycle. Hence the
screen draws no current over a portion of the
modulating cycle, and over the major portion
of the cycle when the screen does draw current, it presents approximately a square -law
impedance.
Circuits for
ScreenGrid
Laboratory analysis of
a
large
number of circuits for accomModulation
plishing screen modulation has
led to the conclusion that the
audio modulating voltage must be obtained
from a low- impedance source if low- distortion modulation is to be obtained. Figure 4
shows a group of sketches of the modulation
envelope obtained with various types of modulators and also with insufficient antenna coupling. The result of this laboratory work led
to the conclusion that the cathode -follower
modulator of the basic circuit shown in figure
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288
Amplitude Modulation
THE
RADIO
ENVELOPE OBTAINED WITH
INSUFFICIENT ANTENNA
COUPLING
+MOO.
-50
+5.0.
V.
APPROX.
O
Figure
4
SCREEN -MODULATION CIRCUITS
Three common screen modulation circuits are illustrated above. All three circuits
are capable of giving intelligible voice modulation although the waveform distortion
in the circuits of (A) and (B) is likely to be rather severe. The arrangement at (A)
is often called "clamp tube" screen modulation; by returning the grid leak on the
clomp tube to ground the circuit will give controlled- carrier screen modulation. This
circuit has the advantage that it is simple and is well suited to use in mobile transmitters. (B) is an arrangement using a transformer coupled modulator, and offers no
particular advantages. The arrangement at (C) is capable of giving good modulation
linearity due to the low impedance of the cathode-follower modulator. However, due
to the relatively low heater-cathode ratings on tubes suited for use as the modulator, a separate heater supply for the modulator tube normally is required. This limitation makes application of the circuit to the mobile transmitter a special problem,
since an isolated heater supply normally is not available. Shown at (D) as an assistance in the tuning of a screen -modulated transmitter (or any efficiency -modulated
transmitter for that matter) is the type of modulation envelope which results when
loading to the modulated stage is insufficient.
is capable of giving good -quality screen grid modulation, and in addition the circuit
provides convenient adjustments for the carrier level and the output level on negative
modulation peaks. This latter control, P2 in
figure 5, allows the amplifier to be adjusted
in such a manner that negative -peak clipping
cannot take place, yet the negative modulation
peaks may be adjusted to a level just above
that at which sideband splatter will occur.
5
The Cathode Follower Modulator
The cathode follower is
ideally suited for use as
the modulator for a screen-
grid stage since it acts as a relatively low impedance source of modulating voltage for
the screen -grid circuit. In addition the cathode follower modulator allows the supply voltage
both for the modulator and for the screen grid
of the modulated tube to be obtained from the
high -voltage supply for the plate of the screen grid tube or beam tetrode. In the usual case
the plate supply for the cathode follower, and
hence for the screen grid of the modulated
tube, may be taken from the bleeder on the
high- voltage power supply. A tap on the bleeder
may be used, or two resistors may be connected in series to make up the bleeder, with ap-
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HANDBOOK
Modulation Systems
289
propriate values such that the voltage applied
to the plate of the cathode follower is appropriate for the tube to be modulated. It is important that a bypass capacitor be used from
the plate of the cathode - follower modulator
to ground.
The voltage applied to the plate of the
cathode follower should be about 100 volts
greater than the rated screen voltage for the
tetrode tube as a c -w Class C amplifier. Hence
the cathode -follower plate voltage should be
about 350 volts for an 815, 2E26, or 829B,
about 400 volts for an 807 or 4 -125A, about
500 volts for an 813, and about 600 volts for
a 4 -250A or a 4E27. Then potentiometer P1
in figure 5 should be adjusted until the carrier level screen voltage on the modulated stage
is about one -half the rated screen voltage
specified for the tube as a Class C c -w amplifier. The current taken by the screen of the
modulated tube under carrier conditions will
be about one - fourth the normal screen current
for c -w operation.
The only current taken by the cathode
follower itself will be that which will flow
through the 100,000 -ohm resistor between the
cathode of the 6L6 modulator and the negative supply. The current taken from the bleeder
on the high -voltage supply will be the carrier level screen current of the tube being modulated (which current passes of course through
the cathode follower) plus that current which
will pass through the 100,000 -ohm resistor.
The loading of the modulated stage should
be adjusted until the input to the tube is about
50 per cent greater than the rated plate dissipation of the tube or tubes in the stage. If the
carrier -level screen voltage value is correct
for linear modulation of the stage, the loading
will have to be somewhat greater than that
amount of loading which gives maximum output
from the stage. The stage may then be modulated by applying an audio signal to the grid
of the cathode -follower modulator, while observing the modulated envelope on an oscilloscope.
If good output is being obtained, and the
modulation envelope appears as shown in figure 4C, all is well, except that P2 in figure 5
should be adjusted until negative modulation
peaks, even with excessive modulating signal,
do not cause carrier cutoff with its attendant
sideband splatter. If the envelope appears as
at figure 4D, antenna coupling should be increased while the carrier level is backed down
by potentiometer PI in figure 5 until a set of
adjustments is obtained which will give a satisfactory modulation envelope as shown in
figure 4C.
Changing Bands
After a satisfactory set of adjustments has been obtained,
Figure 5
CATHODE -FOLLOWER
SCREEN -MODULATION CIRCUIT
A
detailed discussion of this circuit, which
also is represented in figure 4C, is given in
the accompanying text.
it is
not difficult to readjust the amplifier for
operation on different bands. Potentiometers
P1 (carrier level), and P2 ( negative peak level)
may be left fixed after a satisfactory adjustment, with the aid of the scope, has once been
found. Then when changing bands it is only
necessary to adjust excitation until the correct
value of grid current is obtained, and then to
adjust antenna coupling until correct plate
current is obtained. Note that the correct plate
current for an efficiency -modulated amplifier
is only slightly less than the out -of- resonance
plate current of the stage. Hence carrier -level
screen voltage must be low so that the out -ofresonance plate current will not be too high,
and relatively heavy antenna coupling must be
used so that the operating plate current will
be near the out -of- resonance value, and so that
the operating input will be slightly greater
than 1.5 times the rated plate dissipation of
the tube or tubes in the stage. Since the carrier
efficiency of the stage will be only 35 to 40
per cent, the tubes will be operating with plate
dissipation of approximately the rated value
without modulation.
Speech Clipping in
The maximum r -f output
of an efficiency -modulated stage is limited
by the maximum possible plate voltage swing
on positive modulation peaks. In the modula lation circuit of figure 5 the minimum output
is limited by the minimum voltage which the
screen will reach on a negative modulation
peak, as set by potentiometer P2 Hence
the
screen -grid- modulated stage, when using
the
modulator of figure 5, acts effectively as a
speech clipper, provided the modulating signal
amplitude is not too much more than that value
the Modulated Stage
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290
THE
Amplitude Modulation
which will accomplish full modulation. With
correct adjustments of the operating conditions
of the stage it can be made to clip positive
and negative modulation peaks symmetrically.
However, the inherent peak clipping ability of
the stage should not be relied upon as a means
of obtaining a large amount of speech compression, since excessive audio distortion and
excessive screen current on the modulated
stage will result.
Characteristics of
Typical Screen
Modulated Stage
a
An
important character -
istic of the screen-modulated stage, when using
the cathode -follower modulator, is that excessive plate voltage on the
modulated stage is not required. In fact, full
output usually may be obtained with the larger
tubes at an operating plate voltage from one half to two- thirds the maximum rated plate
voltage for c -w operation. This desirable condition is the natural result of using a low impedance source of modulating signal for
the stage.
As an example of a typical screen -modulated stage, full output of 75 watts of carrier
may be obtained from an 813 tube operating
with a plate potential of only 1250 volts. No
increase in output from the 813 may be obtained by increasing the plate voltage, since
the tube may be operated with full rated plate
dissipation of 125 watts, with normal plate
efficiency for a screen -modulated stage, 37.5
per cent, at the 1250-volt potential.
The operating conditions of a screen -modulated 813 stage are as follows:
Plate voltage-1250 volts
Plate current -160 ma.
Plate input -200 watts
Grid current -11 ma.
Grid bias -I 10 volts
Carrier screen voltage -190 volts
Carrier screen current -6 ma.
Power output -approx. 75 watts
With full 100 per cent modulation the plate
current decreases about 2 ma. and the screen
current increases about 1 ma.; hence plate,
screen, and grid current remain essentially
constant with modulation. Referring to figure
5, which was the circuit used as modulator
for the 813, (El) measured plus 155 volts, (E2)
measured -50 volts, (E3) measured plus 190
volts, (Et) measured plus 500 volts, and the
r.m.s. swing at (E5) for full modulation measured 210 volts, which represents a peak swing
of about 296 volts. Due to the high positive
voltage, and the large audio swing, on the
cathode of the 6L6 (triode connected) modulator tube, it is important that the heater of
of this tube be fed from a separate filament
RADIO
transformer or filament winding. Note also that
the operating plate -to- cathode voltage on the
6L6 modulator tube does not exceed the 360 volt rating of the tube, since the operating
potential of the cathode is considerably above
ground potential.
Still another form of efficiency modulation may be
obtained by applying the
audio modulating signal to the suppressor grid
of a pentode Class C r -f amplifier. Basically,
suppressor -grid modulation operates in the
same general manner as other forms of efficiency modulation; carrier plate circuit efficiency is about 35 per cent, and antenna coupling must be rather tight. However, suppressor grid modulation has one sizeable disadvantage,
in addition to the fact that pentode tubes are
not nearly so widely used as beam tetrodes
which of course do not have the suppressor
element. This disadvantage is that the screen grid current to a suppressor -grid modulated
amplifier is rather high. The high screen current is a natural consequence of the rather high
negative bias on the suppressor grid, which
reduces the plate- voltage swing and plate current with a resulting increase in the screen
current.
In tuning a suppressor -grid modulated amplifier, the grid bias, grid current, screen voltage, and plate voltage are about the same as
for Class C c -w operation of the stage. But
the suppressor grid is biased negatively to a
value which reduces the plate- circuit efficiency to about one -half the maximum obtainable
from the particular amplifier, with antenna
coupling adjusted until the plate input is about
1.5 times the rated plate dissipation of the
stage. It is important that the input to the
screen grid be measured to make sure that the
rated screen dissipation of the tube is not
being exceeded. Then the audio signal is applied to the suppressor grid. In the normal
application the audio voltage swing on the
suppressor will be somewhat greater than the
negative bias on the element. Hence suppressor -grid current will flow on modulation
peaks, so that the source of audio signal voltage must have good regulation. Tubes suitable
for suppressor -grid modulation are: 2E22,
837, 4E27/8001, 5 -125, 804 and 803. A typical suppressor -grid modulated amplifier is
illustrated in figure 6.
Suppressor -Grid
Modulation
15 -4
Input Modulation
Systems
Constant efficiency variable -input modulation systems operate by virtue of the addition
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HANDBOOK
Plate Modulation
CARRIER
OUTPUT
4E27
'33w
R.F INPUT
-
IG=
AIA
ISG'
44 M
-130
6J5
V.
2.1 STEPUP
IP=)OMA.
+1500
V.
PEAK SWING FOR FULL
MODULATION = 210 V.
A.F INPUT
+300 V
-210
Figure
V.
6
AMPLIFIER WITH SUPPRESSOR -GRID
MODULATION
Recommended operating conditions for linear suppressor-grid modulation of a 4E27/
2578/8001 stage are given on the drawing.
of external power to the modulated stage to
effect the modulation. There are two general
classifications that come under this heading;
those systems in which the additional power
is supplied as audio frequency energy from a
modulator, usually called plate modulation
systems, and those systems in which the additional power to effect modulation is supplied
as direct current from the plate supply.
Under the former classification comes Heising modulation (probably the oldest type of
modulation to be applied to a continuous carrier), Class B plate modulation, and series
modulation. These types of plate modulation
are by far the easiest to get into operation,
and they give a very good ratio of power input
to the modulated stage to power output; 65 to
80 per cent efficiency is the general rule. It
is for these two important reasons that these
modulation systems, particularly Class B plate
modulation, are at present the most popular
for communications work.
Modulation systems coming under the second classification are of comparatively recent
development but have been widely applied to
broadcast work. There are quite a few systems
in this class. Two of the more widely used
are the Doherty linear amplifier, and the Ter man- Woodyard high- efficiency grid- modulated
amplifier. Both systems operate by virtue of
a carrier amplifier and a peak amplifier connected together by electrical quarter -wave
lines. They will be described later in this
section.
Plate Modulation
Plate modulation is the application of the audio power
291
to the plate circuit of an r -f amplifier. The r -f
amplifier must be operated Class C for this
type of modulation in order to obtain a radio frequency output which changes in exact accordance with the variation in plate voltage.
The r -f ampli fier is 100 per cent modulated
when the peak a -c voltage from the modulator
is equal to the d.c. voltage applied to the r -f
tube. The positive peaks of audio voltage increase the instantaneous plate voltage on the
r -f tube to twice the .1c value, and the negative peaks reduce the voltage to zero.
The instantaneous plate current to the r -f
stage also varies in accordance with the modulating voltage. The peak alternating current
in the output of a modulator must be equal to
the d -c plate current of the Class C r -f stage
at the point of 100 per cent modulation. This
combination of change in audio voltage and
current can be most easily referred to in terms
of audio power in watts.
In a sinusoidally modulated wave, the antenna current increases approximately 22 per
cent for 100 per cent modulation with a pure
tone input; an r -f meter in the antenna circuit
indicates this increase in antenna current.
The average power of the r -f wave increases
50 per cent for 100 per cent modulation, the
efficiency remaining constant.
This indicates that in a plate- modulated
radiotelephone transmitter, the audio- frequency
channel must supply this additional 50 per
cent increase in average power for sine -wave
modulation. If the power input to the modulated stage is 100 watts, for example, the
average power will increase to 150 watts at
100 per cent modulation, and this additional
50 watts of power must be supplied by the
modulator when plate modulation is used. The
actual antenna power is a constant percentage
of the total value of input power.
One of the advantages of plate (or power)
modulation is the ease with which proper adjustments can be made in the transmitter. Also.
there is less plate loss in the r -f amplifier for
a given value of carrier power than with other
forms of modulation because the plate efficiency is higher.
By properly matching the plate impedance
of the r -f tube to the output of the modulator,
the ratio of voltage and current swing to d -c
voltage and current is automatically obtained.
The modulator should have a peak voltage
output equal to the average d -c plate voltage
on the modulated stage. The modulator should
also have a peak power output equal to the
d -c plate input power to the modulated stage.
The average power output of the modulator will
depend upon the type of waveform. If the amplifier is being Heising modulated by a Class
A stage, the modulator must have an average
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292
CLASS
MODULATED CLASS C
R.
RADIO
THE
Amplitude Modulation
C
AMPLIFIER
f. AMPLIFIER
CLASS
IS
MODULATOR
+9
Figure
7
HEISING PLATE MODULATION
This type of modulation was the first form
of plate modulation. It is sometimes known
as "constant current" modulation. Because
of the effective 1:1 ratio of the coupling
choke, it is impossible to obtain 100 per cent
modulation unless the plate voltage to the
modulated stage is dropped slightly by resistor R. The capacitor C merely byp
the audio around R, so that the full a-f output voltage of the modulator is impressed
on the Class C stage.
power output capability of one -half the input
to the Class C stage. If the modulator is a
Class B audio amplifier, the average power
required of it may vary from one -quarter to more
than one -half the Class C input depending
upon the waveform. However, the peak power
output of any modulator must be equal to the
Class C input to be modulated.
Heising modulation is the oldest
system of plate modulation, and
usually consists of a Class A
audio amplifier coupled to the r -f amplifier by
means of a modulation choke coil, as shown
in figure 7.
The d.c. plate voltage and plate current in
the r-f amplifier must be adjusted to a value
which will cause the plate impedance to match
Heising
Modulation
the output of the modulator, since the modulation choke gives a 1 -to -1 coupling ratio. A
series resistor, by- passed for audio frequencies by means of a capacitor, must be connected in series with the plate of the r -f amplifier
to obtain modulation up to 100 per cent. The
peak output voltage of a Class A amplifier
does not reach a value equal to the d -c voltage
applied to the amplifier and, consequently,
the d -c plate voltage impressed across the
r -f tube must be reduced to a value equal to
MOD.
Figure
+5
R F.
.13
8
PLATE MODULATION
This type of modulation is the most flexible
in that the loading adjustment can be made
in a short period of time and without elaborate test equipment after a change in operating frequency of the Class C amplifier has
CLASS
B
been made.
if
available a -c peak voltage
100% modulation is to be obtained.
A higher degree of distortion can be tolerthe maximum
ated in low -power emergency phone transmitters
which use a pentode modulator tube, and the
series resistor and by -pass capacitor are
usually omitted in such transmitters.
High -level Class B plate
modulation is the least expensive method of plate
modulation. Figure 8 shows a conventional
Class B plate -modulated Class C amplifier.
The statement that the modulator output
power must be one -half the Class C input for
100 per cent modulation is correct only if the
waveform of the modulating power is a sine
wove. Where the modulator waveform is unclipped speech, the average modulator power
for 100 per cent modulation is considerably
less than one -half the Class C input.
Class B
Plata Modulation
It has been determined experimentally that the ratio
of peak to average power
in a speech waveform is approximately 4 to 1
as contrasted to a ratio of 2 to 1 in a sine
wave. This is due to the high harmonic content of such a waveform, and to the fact that
Power Relations in
Speech Waveforms
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HANDBOOK
Plate Modulation
this high harmonic content manifests itself by
making the wave unsymmetrical and causing
sharp peaks or "fingers" of high energy content to appear. Thus for unclipped speech, the
average modulator plate current, plate dissipation, and power output are approximately
one -half the sine wave values for a given peak
output power.
Both peak power and average power are
necessarily associated with waveform. Peak
power is just what the name implies; the power
at the peak of a wave. Peak power, although
of the utmost importance in modulation, is of
no great significance in a -c power work, except insofar as the average power may be determined from the peak value of a known wave
form.
There is no time element implied in the
definition of peak power; peak power may be
instantaneous -and for this reason average
power, which is definitely associated with
time, is the important factor in plate dissipation. It is possible that the peak power of a
given waveform be several times the average
value; for a sine wave, the peak power is twice
the average value, and for unclipped speech
the peak power is approximately four times
the average value. For 100 per cent modulation, the peak (instantaneous) audio power
must equal the Class C input, although the
average power for this value of peak varies
widely depending upon the modulator waveform, being greater than 50 per cent for speech
that has been clipped and filtered, 50 per cent
for a sine wave, and about 25 per cent for typical unclipped speech tones.
Modulation
Transformer
Calculations
The
modulation
transformer is
a device for matching the load
impedance of the Class C amplifier to the recommended load
impedance of the Class B modulator tubes.
Modulation transformers intended for communications work are usually designed to
carry the Class C plate current through their
secondary windings, as shown in figure 8.
The manufacturer's ratings should be consulted to insure that the d-c plate current
passed through the secondary winding does
not exceed the maximum rating.
A detailed discussion of the method of
making modulation transformer calculations
has been given in Chapter Six. However, to
emphasize the method of making the calculation, an additional example will be given.
Suppose we take the case of a Class C amplifier operating at a plate voltage of 2000
with 225 ma. of plate current. This amplifier
would present a load resistance of 2000 divided by 0.225 amperes or 8888 ohms. The plate
power input would be 2000 times 0.225 or 450
watts. By reference to Chapter Six we see that
293
a pair of 811 tubes operating at 1500 plate
volts will deliver 225 watts of audio output.
The plate -to -plate load resistance for these
tubes under the specified operating conditions
is 18,000 ohms. Hence our problem is to match
the Class C amplifier load resistance of 8888
ohms to the 18,000 -ohm load resistance required by the modulator tubes.
A 200 -to -300 watt modulation transformer
will be required for the job. If the taps on the
transformer are given in terms of impedances
it will only be necessary to connect the secondary for 8888 ohms (or a value approximately
equal to this such as 9000 ohms) and the primary for 18,000 ohms. If it is necessary to
determine the proper turns ratio required of the
transformer it can be determined in the following manner. The square root of the impedance
ratio is equal to the turns ratio, hence:
8888
18000
=
V 0.494
=
0.703
The transformer must have a turns ratio of
approximately 1- to -0.7 step down, total primary to total secondary. The greater number
of turns always goes with the higher impedance, and vice versa.
Plate- andScreen
Modulation
only the plate of a
screen -grid tube is modulated, it is impossible to obtain high -percentage linear modulation under
ordinary conditions. The plate current of such
a stage is not linear with plate voltage. However, if the screen is modulated simultaneously
with the plate, the instantaneous screen voltage drops in proportion to the drop in the plate
voltage, and linear modulation can then be obtained. Four satisfactory circuits for accomplishing combined plate and screen modulaWhen
tion are shown in figure 9.
The screen r -f by -pass capacitor C2 should
not have a greater value than 0.005 µfd., preferably not larger than 0.001 tad. It should be
large enough to bypass effectively all r -f voltage without short- circuiting high- frequency
audio voltages. The plate by -pass capacitor
can be of any value from 0.002 µfd. to 0.005
µfd. The screen -dropping resistor, 111. should
reduce the applied high voltage to the value
specified for operating the particular tube in
the circuit. Capacitor C1 is seldom required
yet some tubes may require this capacitor in
order to keep C2 from attenuating the high frequencies. Different values between .0002 and
.002 µfd. should be tried for best results.
Figure 9C shows another method which uses
a third winding on the modulation transformer,
through which the screen -grid is connected to
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294
RADIO
THE
Amplitude Modulation
E
3
3
B+ S.G.
Figure
B+
9
PLATE MODULATION OF A BEAM TETRODE OR SCREEN -GRID TUBE
These alternative arrangements for plate modulation of tetrodes or pentodes are discussed in detail in the text. The arrangements shown at (B) or (D) are recommended
for most applications.
a low- voltage power supply. The ratio of turns
between the two output windings depends upon
the type of screen -grid tube which is being
modulated. Normally it will be such that the
screen voltage is being modulated 60 per cent
when the plate voltage is receiving 100 per
cent modulation.
If the screen voltage is derived from a dropping resistor ( not a divider) that is bypassed
for r.f. but not a.f., it is possible to secure
quite good modulation by applying modulation
only to the plate. Under these conditions, the
screen tends to modulate itself, the screen
voltage varying over the audio cycle as a result of the screen impedance increasing with
plate voltage, and decreasing with a decrease
in plate voltage. This circuit arrangement is
illustrated in figure 9B.
A similar application of this principle is
shown in figure 9D. In this case the screen
voltage is fed directly from a low- voltage supply of the proper potential through a choke L.
A conventional filter choke having an inductance from 10 to 20 henries will be satisfactory for L.
To afford protection of the tube when plate
voltage is not applied but screen voltage is
supplied from the exciter power supply, when
using the arrangement of figure 9D, a resistor
of 3000 to 10,000 ohms can be connected in
series with the choke L. In this case the screen
supply voltage should be at least 1%Z times as
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HANDBOOK
Cathode Modulation
much as is required tor actual screen
voltage,
and the value of resistor is chosen such that
with normal screen current the drop through
the resistor and choke will be such that normal screen voltage will be applied to the tube.
When the plate voltage is removed the screen
current will increase greatly and the drop
through resistor R will increase to such a
value that the screen voltage will be lowered
to the point where the screen dissipation on
the tube will not be exceeded. However, the
supply voltage and value of resistor R must
be chosen carefully so that the maximum rated
screen dissipation cannot be exceeded. The
maximum possible screen dissipation using
this arrangement is equal to: W = E' /4R where
E is the screen supply voltage and R is the
combined resistance of the resistor in figure
9D and the d -c resistance of the choke L. It
is wise, when using this arrangement to check,
using the above formula, to see that the value
of W' obtained is less than the maximum rated
screen dissipation of the tube or tubes used
in the modulated stage. This same system can
of course also be used in figuring the screen
supply circuit of a pentode or tetrode amplifier stage where modulation is not to be
applied.
The modulation transformer for plate -andscreen- modulation, when utilizing a dropping
resistor as shown in figure 9A, is similar to
the type of transformer used for any plate
modulated phone. The combined screen and
plate current is divided into the plate voltage
in order to obtain the Class C amplifier load
impedance. The peak audio power required to
obtain 100 per cent modulation is equal to the
d-c power input to the screen, screen resistor,
and plate of the modulated r -f stage.
15 -5
Cathode Modulation
Cathode modulation offers a workable compromise between the good plate efficiency but
expensive modulator of high -level plate modulation, and the poor plate efficiency but inexpensive modulator of grid modulation. Cathode
modulation consists essentially of an admixture of the two.
The efficiency of the average well- designed
plate -modulated transmitter is in the vicinity
of 75 to 80 per cent, with a compromise perhaps at 77.5 per cent. On the other hand, the
efficiency of a good grid-modulated transmitter
may run from 28 to maybe 40 per cent, with
the average falling at about 34 per cent. Now
since cathode modulation consists of simultaneous grid and plate modulation, in phase
with each other, we can theoretically obtain
any efficiency from about 34 to 77.5 per cent
from our cathode -modulated stage, depending
295
upon the relative percentages of grid and
plate modulation.
Since the system is a compromise between
the two fundamental modulation arrangements,
a value of efficiency approximately half way
between the two would seem to be the best
compromise. Experience has proved this to be
the case. A compromise efficiency of about
56.5 per cent, roughly half way between the
two limits, has proved to be optimum. Calculation has shown that this value of efficiency can be obtained from a cathode -modulated amplifier when the audio- frequency modulating power is approximately 20 per cent of
the d-c input to the cathode -modulated stage.
Series cathode modulation is
ideally suited as an economiModulator
cal modulating arrangement
for a high -power triode c -w
transmitter. The modulator can be constructed
quite compactly and for a minimum component
cost since no power supply is required for it.
When it is desired to change over from c -w to
'phone, it is only necessary to cut the series
modulator into the cathode return circuit of the
c -w amplifier stage. The plate voltage for the
modulator tubes and for the speech amplifier
is taken from the cathode voltage drop of the
modulated stage across the modulator unit.
Figure 10 shows the circuit of such a modulator, designed to cathode modulate a Class C
amplifier using push -pull 810 tubes, running
at a supply voltage of 2500, and with a plate
input of 660 watts. The modulated stage runs
at about 50% efficiency, giving a power output
of nearly 350 watts, fully modulated. The voltage drop across the cathode modulator is 400
volts, allowing a net plate to cathode voltage
of 2100 volts on the final amplifier. The plate
current of the 810's should be about 330 ma.,
and the grid current should be approximately
40 ma., making the total cathode current of the
modulated stage 370 ma. Four parallel 6L6
modulator tubes can pass this amount of plate
current without difficulty. It must be remembered that the voltage drop across the cathode
modulator is also the cathode bias of the modulated stage. In most cases, no extra grid bias
is necessary. If a bias supply is used for c -w
operation, it may be removed for cathode modulation, as shown in figure 11. With low-mu
triodes, some extra grid bias (over and above
that amount supplied by the cathode modulator)
may be needed to achieve proper linearity of
the modulated stage. In any case, proper operation of a cathode modulated stage should be
determined by examining the modulated output
waveform of the stage on an oscilloscope.
An Economical
Series Cathode
Excitation
r-f driver for a cathode -modulated stage should have about
The
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296
THE
Amplitude Modulation
RADIO
TO CATHODE
MODULATED
STAGE
6L6
6L6
6AU6
6AU6
6
6 L6
L6
500K
.002
T
10 K
l
W
°T
CAUTION
ALL RESISTORS 0.5 wArr (INCEST
OTHERWISE NOTED
ALL CAPACITORS IN LIP UNLESS
OTHERWISE NOTED.
Figure
-
FILAMENTS OF OL° rueES MUST SE Al OPERATING
TEMPERATURE BEFORE PLATE VOLTAGE IS APPLIED
TO MODULATED AMPLIFIER.
10
SERIES CATHODE MODULATOR FOR A HIGH -POWERED TRIODE
R -F
AMPLIFIER
the same power output capabilities as would
be required to drive a c -w amplifier to the same
input as it is desired to drive the cathode modulated stage. However, some form of excitation control should be available since the
amount of excitation power has a direct bearing
on the linearity of a cathode -modulated amplifier stage. If link coupling is used between
the driver and the modulated stage, variation
in the amount of link coupling will afford
apple excitation variation. If much less than
40% plate modulation is employed, the stage
begins to resemble a grid -bias modulated
stage, and the necessity for good r -f regulation will apply.
Cathode modulation has
not proved too satisfactory for use with beam
tetrode tubes. This is a result of the small
excitation and grid swing requirements for
such tubes, plus the fact that some means for
holding the screen voltage at the potential of
the cathode as far as audio is concerned is
usually necessary. Because of these factors,
cathode modulation is not recommended for
use with tetrode r -f amplifiers.
Cathode Modulation
of Tetrodes
15 -6
The Doherty and the
Terman- Woodyard
Modulated Amplifiers
These two amplifiers will be described together since they operate upon very similar
principles. Figure 12 shows a greatly simplified schematic diagram of the operation of both
types. Both systems operate by virtue of a carrier tube (V, in both figures 12 and 13) which
supplies the unmodulated carrier, and whose
output is reduced to supply negative peaks,
and a peak tube (V2) whose function is to
supply approximately half the positive peak
of the modulation cycle and whose additional
function is to lower the load impedance on the
carrier tube so that it will be able to supply
the other half of the positive peak of the modulation cycle.
The peak tube is enabled to increase the
output of the carrier tube by virtue of an impedance inverting line between the plate circuits of the two tubes. This line is designed
to have a characteristic impedance of one -half
the value of load into which the carrier tube
operates under the carrier conditions. Then a
load of one -half the characteristic impedance
of the quarter -wave line is coupled into the
output. By experience with quarter -wave lines
in antenna -matching circuits we know that
such a line will vary the impedance at one
end of the line in such a manner that the geometric mean between the two terminal impedances will be equal to the characteristic impedance of the line. Thus, if we have a value
of load of one -half the characteristic impedance of the line at one end, the other end of
the line will present a value of Juice the characteristic impedance of the lines to the carrier tube V,.
This is the situation that exists under the
carrier conditions when the peak tube merely
floats across the load end of the line and contributes no power. Then as a positive peak of
modulation comes along, the peak tube starts
to contribute power to the load until at the
peak of the modulation cycle it is contributing
enough power so that the impedance at the
load end of the line is equal to R, instead of
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HANDBOOK
R
Doherty Amplifier
F.
AMPLIFIER
vi
297
ELECTRICAL 5/4
(LINE ZO'R
040
LOAD
BIAS SUPPLY
FOR C. W.
MIC
BAIA!
BAUE
PHONE
Figure 12
DIAGRAMMATIC REPRESENTATION OF
THE DOHERTY LINEAR
PHONE
-ELE'S
CATHODE
MODULATOR
Figure 11
MODULATOR INSTALLATION
SHOWING PHONE -C.W. TRANSFER
SWITCH
CATHODE
desirable phase shift of 90° between the plate
circuits of the carrier and peak tubes, an equal
and opposite phase shift must be introduced in
the exciting voltage to the grid circuits of the
two tubes so that the resultant output in the
plate circuit will be in phase. This additional
phase shift has been indicated in figure 12 and
a method of obtaining it has been shown in
figure 13.
The difference between
the Doherty linear amplifier and the TermanGrid Modulator
Woodyard grid -modulated
amplifier is the same as the difference between
any linear and grid -modulated stages.Modulated
r.f.is applied to the grid circuit of the Doherty
linear amplifier with the carrier tube biased to
cutoff and the peak tube biased to the point
where it draws substantially zero plate current
at the carrier condition.
Comparison Between
Linear and
the R/2 that is presented under the carrier
conditions. This is true because at a positive
modulation peak (since it is delivering full
power) the peak tube subtracts a negative
resistance of R/2 from the load end of the
line.
Now, since under the peak condition of modulation the load end of the line is terminated
in R ohms instead of R /2, the impedance at
the carrier -tube will be reduced from 2R ohms
to R ohms. This again is due to the impedance
inverting action of the line. Since the load resistance on the carrier tube has been reduced
to half the carrier value, its output at the peak
of the modulation cycle will be doubled. Thus
we have the necessary condition for a 100
per cent modulation peak; the amplifier will
deliver four times as much power as it does
under the carrier conditions.
On negative modulation peaks the peak tube
does not contribute; the output of the carrier
tube is reduced until on a 100 per cent negative peak its output is zero.
While an electrical quarter wave line (consisting of a pi
Line
network with the inductance
and capacitance units having
a reactance equal to the characteristic impedance of the line) does have the desired impedance- inverting effect, it also has the undesirable effect of introducing a 90° phase
shift across such a line. If the shunt elements
are capacitances, the phase shift across the
line lags by 90 °; if they are inductances, the
phase shift leads by 90 °. Since there is an unThe Electrical
Quorter -Wave
In the Terman -Woodyard grid-modulated amthe carrier tube runs Class C with comparatively high bias and high plate efficiency,
while the peak tube again is biased so that it
draws almost no plate current. Unmodulated
r.f. is applied to the grid circuits of the two
tubes and the modulating voltage is inserted
in series with the fixed bias voltages. From
one -half to two -thirds as much audio voltage
is required at the grid of the peak tube as is
required at the grid of the carrier tube.
plifier
The resting carrier efficiency of
the grid- modulated amplifier may
run as high as is obtainable in
any Class C stage, 80 per cent or better. The
resting carrier efficiency of the linear will be
about as good as is obtainable in any Class
13
amplifier, 60 to 70 per cent. The overall
efficiency of the bias -modulated amplifier at
100 per cent modulation will run about 75 per
cent; of the linear, about 60 per cent.
In figure 13 the plate tank circuits are detuned enough to give an effect equivalent to
the shunt elements of the quarter -wave "line"
of figure 12. At resonance, the coils L, and
L2 in the grid circuits of the two tubes have
Operating
Efficiencies
www.americanradiohistory.com
298
THE
Amplitude Modulation
NC
Q
LI
ó
EXCITATION
BIAS
r
a
L30
d
V.I-1_
other undesirable features which make their
use impracticable alongside the more conventional modulation systems. Nearly all these
circuits have been published in the 1.R.E.
Proceedings and the interested reader can refer to them in back copies of that journal.
Ci
15 -7
TO
O
T
ANT.
Tc3
2
Figure
13
SIMPLIFIED SCHEMATIC OF A
"HIGH EFFICIENCY" AMPLIFIER
The basic system, comprising a "carrier"
tube and a "peak" tube interconnected by
lumped -constant quarter -wave lines, is the
some for either grid-bias modulation or for
use as a linear amplifier of a modulated
wave.
each an inductive reactance equal to the capacitive reactance of the capacitor C1, Thus
we have the effect of a pi network consisting
of shunt inductances and series capacitance.
In the plate circuit we want a phase shift of
the same magnitude but in the opposite direction; so our series element is the inductance
L3 whose reactance is equal to the characteristic impedance desired of the network. Then
the plate tank capacitors of the two tubes C2
and C3 are increased an amount past resonance, so that they have a capacitive reactance
equal to the inductive reactance of the coil L3.
It is quite important that there be no coupling
between the inductors.
Although both these types of amplifiers are
highly efficient and require no high -level audio
equipment, they are difficult to adjust- particularly so on the higher frequencies -and it
would be an extremely difficult problem to design a multiband transmitter employing the
circuit. However, the grid -bias modulation system has advantages for the high -power transmitter which will be operated on a single fre-
quency band.
Other High- Efficiency
Modulation Systems
Many other high- efficiency modulation systems
have been described since
about 1936. The majority of these, however
have received little application either by commercial interests or by amateurs. In most cases
the circuits are difficult to adjust, or they have
RADIO
Speech Clipping
Speech waveforms are characterized by frequently recurring high -intensity peaks of very
short duration. These peaks will cause over modulation if the average level of modulation
on loud syllables exceeds approximately 30
per cent. Careful checking into the nature of
speech sounds has revealed that these high intensity peaks are due primarily to the vowel
sounds. Further research has revealed that the
vowel sounds add little to intelligibility, the
major contribution to intelligibility coming
from the consonant sounds such as v, b, k, s,
t, and 1. Measurements have shown that the
power contained in these consonant sounds
may be down 30 db or more from the energy in
the vowel sounds in the same speech passage.
Obviously, then, if we can increase the relative energy content of the consonant sounds
with respect to the vowel sounds it will be
possible to understand a signal modulated with
such a waveform in the presence of a much
higher level of background noise and interference. Experiment has shown that it is possible to accomplish this desirable result simply by cutting off or clipping the high- intensity
peaks and thus building up in a relative manner the effective level of the weaker sounds.
Such clipping theotetically can be accomplished simply by increasing the gain of the
speech amplifier until the average level of
modulation on loud syllables approaches 90
per cent. This is equivalent to increasing the
speech power of the consonant sounds by about
10 times or, conversely, we can say that 10 db
of clipping has been applied to the voice wave.
However, the clipping when accomplished in
this manner will produce higher order side bands known as "splatter," and the transmitted
signal would occupy a relatively tremendous
slice of spectrum. So another method of accomplishing the desirable effects of clipping must
be employed.
A considerable reduction in the amount of
splatter caused by a moderate increase in the
gain of the speech amplifier can be obtained
by poling the signal from the speech amplifier
to the transmitter such that the high- intensity
peaks occur on upward or positive modulation.
Overloading on positive modulation peaks produces less splatter than the negative -peak
clipping which occurs with overloading on the
www.americanradiohistory.com
Speech
HANDBOOK
Clipping
299
Figure 14
SPEECH -WAVEFORM AMPLITUDE
MODULATION
Showing the effect of using the proper polarity of a speech wave for
modulating a transmitter. (A) shows
the effect of proper speech polarity
on a transmitter having an upward
modulation capability of greater
than 100 per cent. (B) shows the
effect of using proper speech polarity on a transmitter having an upward modulation capability of only
100 per cent. Both these conditions
will give a clean signal without
objectionable splatter. (C) shows
the effect of the use of improper
speech polarity. This condition will
cause serious splatter due to negative -peak clipping in the modulated amplifier stage.
1001b NEG MODULATION
_100
Q
AVERAGE LEVEL
100 % NEG. MODULATION
100
%
POS. MODULATION
AVERAGE LEVEL
r
NEGATIVE
PEAR CLIPPING
negative peaks of modulation. This aspect of
the problem has been discussed in more detail
in the section on Speech Waveform Dissymmetry
earlier in this chapter. The effect of feeding
the proper speech polarity from the speech amplifier is shown in figure 14.
A much more desirable and effective method
of obtaining speech clipping is actually to employ a clipper circuit in the earlier stages of
the speech amplifier, and then to filter out the
objectionable distortion components by means
of a sharp low -pass filter having a cut-off frequency of approximately 3000 cycles. Tests on
clipper -filter speech systems have shown that
6 db of clipping on voice is just noticeable,
12 db of clipping is quite acceptable, and
values of clipping from 20 to 25 db are tolerable under such conditions that a high degree
of clipping is necessary to get through heavy
QRM or QRN. A signal with 12 db of clipping
doesn't sound quite natural but it is not unpleasant to listen to and is much more readable than an unclipped signal in the presence
of strong interference.
The use of a clipper- filter in the speech amplifier, to be completely effective, requires
that phase shift between the clipper- filter
stage and the final modulated amplifier be kept
%b POS. MODULAT I
100
%b
NEG. MODULATION
1
a minimum. However, if there is phase shift
after the clipper- filter the system does not
completely break down. The presence of phase
shift merely requires that the audio gain following the clipper- filter be reduced to the point
where the cant applied to the clipped speech
waves still cannot cause overmodulation. This
effect is illustrated in figures 15 and 16.
The cant appearing on the tops of the square
waves leaving the clipper -filter centers about
the clipping level. Hence, as the frequency
being passed through the system is lowered,
the amount by which the peak of the canted
wave exceeds the clipping level is increased.
to
In a normal transmitter having a
moderate amount of phase shift
the cant applied to the tops of
the waves will cause overmodulation on frequencies below those for which the gain following the clipper -filter has been adjusted unless remedial steps have been taken. The following steps are advised:
(1) Introduce bass suppression into the speech
amplifier ahead of the clipper- filter.
Phase Shift
Correction
(2)
improve the low- frequency response characteristic insofar as it is possible in the
www.americanradiohistory.com
300
THE
Amplitude Modulation
RADIO
POSITIVE CLIPPING LEVEL
AVERAGE LEVEL
IttGATIVE ÇLIPPMGJ-41F,L
INCOMING SPEECH WAVE
POSITIVE CLIPPING LEVEL
AVERAGE LEVEL
NEGATIVE CLIPPING LEVEL
CLIPPED AND FILTERED SPEECH WAVE
_100%
POSITIVE MODULATION
70% POSITIVEMOOUÇATIQN
AVERAGE LEVEL
Figure 15
ACTION OF A CLIPPER -FILTER
ON A SPEECH WAVE
The drawing (A) shows the incoming speech wave before it reaches
the clipper stage. (B) shows the
output of the clipper- filter, illustrating the manner in which the
peaks are clipped and then the
sharp edges of the clipped wave
removed by the filter. (C) shows
the effect of p hase shift In the
stages following the clipper- filter.
(C) also shows the manner in which
the transmitter may be adjusted for
100 per cent modulation of' the
"canted" peaks of the wave, the
sloping top of the wave reaching
about 70 per cent modulation.
70 % NEGATIVE MODULATION
100% NEGATIVE MODULATION
MODULATED WAVE AFTER UNDERGOING PHASE SHIFT
stages following the clipper -filter. Feeding the plate current to the final amplifier
through a choke rather than through the
secondary of the modulation transformer
will help materially.
Even with the normal amount of improvement
which can be attained through the steps mentioned above there will still be an amount of
wave cant which must be compensated in some
manner. This compensation can be done in
either of two ways. The first and simpler way
is as follows:
(1)
Adjust the speech gain ahead of the clipper- filter until with normal talking into
the microphone the distortion being introduced by the clipper -filter circuit is quite
apparent but not objectionable. This amount
of distortion will be apparent to the normal
listener when 10 to 15 db of clipping is
taking place.
2)
Tune a selective communications receiver
about 15 kc. to one side or the other of the
frequency being transmitted. Use a short
antenna or no antenna at all on the receiver so that the transmitter is not blocking the receiver.
(
(3)
Again with the normal talking into the
microphone adjust the gain following the
clipper -filter to the point where the side band splatter is being heard, and then
slightly back off the gain after the clipper- filter until the splatter disappears.
If the phase shift in the transmitter or modulator is not excessive the adjustment procedure given above will allow a clean signal. to be
radiated regardless of any reasonable voice
level being fed into the microphone.
If a cathode -ray oscilloscope is available
the modulated envelope of the transmitter
should be checked with 30 to 70 cycle saw tooth waves on the horizontal axis. If the upper
half of the envelope appears in general the
same as the drawing of figure 15C, all is well
and phase -shift is not excessive. However, if
much more slope appears on the tops of the
waves than is illustrated in this figure, it will
be well to apply the second step in compensation in order to insure that sideband splatter
cannot take place and to afford a still higher
average percentage of modulation. This second
step consists of the addition of a high -level
splatter suppressor such as is illustrated in
figure
17.
www.americanradiohistory.com
HANDBOOK
T
Splatter
n
MODULATOR
301
Suppression
5R4GY, 1616
836
Cz
tit
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111
C4
1/
1
TO
J
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If
3000%
WAVE
FIL. TRANS
INSULATED
PLATE-MODULATED
CLASS-C AMPLIFIER
7500 -10 000 OHMS
LOAD
FOR N.V.
+B MOD.
115
+e R.F. FINAL
V.A.C.
Figure
17
HIGH -LEVEL SPLATTER SUPPRESSOR
This circuit is effective in reducing splatter
caused by negative -peak clipping in the modulated amplifier stage. The use of a two section filter as shown is recommended, although either a single m- derived or a con stant-k section may be used for greater economy. Suitable chokes, along with recommended capacitor values, are available from
several manufacturers.
1000% WAVE
3001.
Figure
WAVE
16
ILLUSTRATING THE EFFECT OF PHASE
SHIFT AND FILTERED WAVES OF DIFFERENT FREQUENCY
Sketch (A) shows the effect of a clipper and
a filter having a cutoff of about 3500 cycles
on o wave of 3000 cycles. Note that no harmonics ore present in the wave so that phase
shift following the clipper -filter will have
no significant effect on the shape of the
wave. (B) and (C) show the effect of phase
shift on waves well below the cutoff frequency of the filter. Note that the "cant" placed
upon the top of the wave causes the peak
value to rise higher and higher above the
clipping level as the frequency is lowered.
It is for this reason that bass suppression
before the clipper stage is desirable. Im-
proved low -frequency response following the
clipper -filter will reduce the phase shift and
therefore the canting of the wave at the lower
voice frequencies.
The use of a high -level splatter suppressor
after a clipper -filter system will afford the result shown in figure 18 since such a device
will not permit the negative -peak clipping
which the wave cant caused by audio -system
phase shift can produce. The high -level splatter suppressor operates by virtue of the fact
that it will not permit the plate voltage on the
modulated amplifier to go completely to zero
regardless of the incoming signal amplitude.
Hence negative -peak clipping with its attendant splatter cannot take place. Such a device
can, of course, also be used in a transmitter
which does not incorporate a clipper- filter system. However, the full increase in average
modulation level without serious distortion,
afforded by the clipper- filter system, will not
be obtained.
A word of caution should be noted at this
time in the case of tetrode final modulated
amplifier stages which afford screen voltage
modulation by virtue of a tap or a separate
winding on the modulation transformer such
as is shown in figure 9C of this chapter. If
such a system of modulation is in use, the
high -level splatter suppressor shown in figure
17 will not operate satisfactorily since negative -peak clipping in the stage can take place
when the screen voltage goes too low.
Clipper Circuits
Two effective low -level clip-
per- filter circuits are shown
in figures 19 and 20. The circuit of figure 19
employs a 6J6 double triode as a clipper, each
half of the 6J6 clipping one side of the impressed waveform. The optimum level at which
the clipping operation begins is set by the
value of the cathode resistor. A maximum of
12 to 14 db of clipping may be used with this
circuit, which means that an extra 12 to 14 db
of speech gain must precede the clipper. For
a peak output of 8 volts from the clipper -filter,
a peak audio signal of about 40 volts must be
impressed upon the clipper input circuit. The
6C4 speech amplifier stage must therefore be
considered as a part of the clipper circuit as
www.americanradiohistory.com
Amplitude Modulation
302
THE
%
100
RADIO
Figure 18
ACTION OF HIGH -LEVEL
POS MODULATION
SPLATTER SUPPRESSOR
high -level splatter suppressor
may be used in a transmitter without a clipper-filter to reduce negative -peak clipping, or such a unit
may be used following a clipper filter to allow a higher average
modulation level by eliminating the
negative -peak clipping which the
wave -cant caused by phase shift
might produce.
A
ZERO AXIS
100
lb
NEG. MODULATION
SPLATTER- CAUSING
NEGATIVE OVERMODULATION PEAK
CUT OFF BY "NIGH -LEVEL
SPLATTER SUPPRESSOR"
the 12 to 14 db loss of gain
incurred in the clipping process. A simple low pass filter made up of a 20 henry a.c. - d.c
replacement type filter choke and two mica
condensers follows the 616 clipper. This filter is designed for a cutoff frequency of about
3500 cycles when operating into a load impedance of % megohm. The output level of 8
volts peak is ample to drive a triode speech
amplifier stage, such as a 6C4 or 6J5.
A 6AL5 double diode series clipper is employed in the circuit of figure 20, and a commercially made low -pass filter is used to give
somewhat better high frequency cutoff characteristics. A double triode is employed as a
speech amplifier ahead of the clipper circuit.
The actual performance of either circuit is
about the same.
To eliminate higher order products that may
be generated in the stages following the clipper- filter, it is wise to follow the modulator
with a high -level filter, as shown in figure 21.
it compensates for
Clipper Adjustment
These
clipper
circuits
have two adjustments:
Adjust Gain and Adjust Clipping. The Adj.
Gain control determines the modulation level
of the transmitter. This control should be set
so that over-modulation of the transmitter is
impossible, regardless of the amount of clipping used. Once the Adj. Gain control has
been roughly set, the Adj. Clip. control may
be used to set the modulation level to any percentage below 100 %. As the modulation level
is decreased, more and more clipping is introduced into the circuit, until a full 12 db of
clipping is used. This means that the Adj.
Gain control may be advanced some 12 db past
the point where the clipping action started.
Clipping action should start at 85% to 90%
modulation when a sine wave is used for circuit adjustment purposes.
Even though we may have cut off
all frequencies above 3000 or 3500
cycles through the use of a filter
system such as is shown in the circuits of figures 19 and 20, higher frequencies may again
be introduced into the modulated wave by
distortion in stages following the speech amplifier. Harmonics of the incoming audio frequencies may be generated in the driver stage
for the modulator; they may be generated in the
plate circuit of the modulator; or they may be
generated by non -linearity in the modulated
High -Level
Filters
amplifier itself.
6J6
6C4
6ÁU6
ADJUST CL/P. A7oULF
ATAL
MIC.
SSK
20
N
jSTANCOR
Cilllj
ADJUST GAIN
AI
4.711
NEXT
GRID
TO
500
11
PEAK OUTPUT APPROX.
/Z OE
!V
MAX W/TN
OF CLIPPING.
5K,
ALL RES /STOPS O.5 WATT UNLESS
OTHERWISE MARKED
ALL CAPACITORS /N {/E UNLESS
OTHERWISE NOTED.
K
w
Y250
v.
Figure
19
CLIPPER FILTER USING 6J6 DOUBLE TRIODE STAGE
www.americanradiohistory.com
HANDBGOK
Splatter
12AX7
6AL5
ADJUST GAIN
Suppression
303
CHICAGO TRANS.
LPF -2 FILTER
TO NEXT
GRID
4
7n
SeK
PEAK OUTPUTAPPROX
5V MAX. WITH 12 DB
OF CLIPPING
ADJUST
SN,IW
00N
CLIP.
ALL RESISTORS 0.5 WATT UNLESS
OTHERWISE MARKED
ALL CAPACITORS IN OF UNLESS
OTHERWISE MARKED.
Figure 20
CLIPPER FILTER USING 6AL5 STAGE
Regardless of the point in the system following the speech amplifier where the high
audio frequencies may be generated, these frequencies can still cause a broad signal to be
transmitted even though all frequencies above
3000 or 3500 cycles have been cut off in the
speech amplifier. The effects of distortion in
the audio system following the speech amplifier can be eliminated quite effectively through
the use of a post- modulator filter. Such a filter
must be used between the modulator plate circuit and the r -f amplifier which is being modulated.
This filter may take three general forms in
a normal case of a Class C amplifier plate modulated by a Class B modulator. The best method is to use a high level low -pass filter as
CLASS
C
AMPLIFIER
shown in figure 21 and discussed previously.
Another method which will give excellent results in some cases and poor results in others,
dependent upon the characteristics of the modulation transformer, is to "build out" the mod-
ulation transformer into a filter section. This
is accomplished as shown in figure 22 by placing mica capacitors of the correct value across
the primary and secondary of the modulation
transformer. The proper values for the capaci-
CLASS
C
STAGE
MODULATOR
MODULATION
TRANSFORMER
B+ MOD.
BI CLASS
C
Figure 22
"BUILDING -OUT" THE MODULATION
Figure 21
ADDITIONAL HIGH -LEVEL LOW -PASS FILTER TO
FOLLOW MODULATOR WHEN A
LOW -LEVEL CLIPPER FILTER IS USED
Suitable choke, along with recommended capacitor values, is available from several
manufacturers.
TRANSFORMER
This expedient utilizes the leakage reactance of the modulation transformer in conjunction with the capacitors shown to make
up a single- section low -pass filter. In order
to determine exact values for CI and C2 plus
C3, it is necessary to use a measurement
setup such as Is shown in figure 23. However, experiment has shown in the case of a
number of commercially available modulation
transformers that a value for Cf of 0.002 -µfd.
and C2 plus C3 of 0.004-µfd. will give satisfactory results.
www.americanradiohistory.com
304
Amplitude Modulation
,
RADIO
THE
1
11111
E%I/NÌ/111
AUDIO
OSCILLATOR
1
I
M
3íMIA11/1I1
4=11N11111
mri 1111111
oazA1/11111
2
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did
Figure 23
TEST SETUP FOR BUILDING -OUT
MODULATION TRANSFORMER
Through the use of a test setup such as is
shown and the method described in the text
it is possible to determine the correct values
for a specified filter characteristic in the
built -out modulation transformer.
tors C1 and C2 must, in the ideal case, be determined by trial and error. Experiment with
a number of modulators has shown, however,
that if a 0.002 pfd. capacitor is used for Cl
and if the sum of C2 and C3 is made 0.004 tad.'
(0.002 pfd. for CZ and 0.002 for C3) the ideal
condition of cutoff above 3000 cycles will be
approached in most cases with the "multiple match" type of modulation transformer.
If it is desired to determine the optimum
values of the capacitors across the transformer
this can be determined in several ways, all of
which require the use of a calibrated audio
oscillator. One way is diagrammed in figure 23.
The series resistors R1 and R2 should each be
equal to V2 the value of the recommended plate to -plate load resistance for the Class B modulator tubes. Resistor R3 should be equal to the
value of load resistance which the Class C
modulated stage will present to the modulator.
The meter V can be any type of a -c voltmeter.
The indicating instrument on the secondary of
the transformer can be either a cathode -ray
oscilloscope or a high -impedance a -c voltmeter of the vacuum -tube or rectifier type.
With a set -up as shown in figure 23 a plot of
output voltage against frequency is made, at
all times keeping the voltage across V constant, using various values of capacitance for
C1 and C2 plus C3, When the proper values of
capacitance have been determined which give
substantially constant output up to about 3000
or 3500 cycles and decreasing output at all frequencies above, high -voltage mica capacitors
can be substituted if receiving types were used
in the tests and the transformer connected to
the modulator and Class C amplifier.
With the transformer reconnected in the
transmitter a check of the modulated -wave
output of the transmitter should be made using
an audio oscillator as signal generator and an
oscilloscope coupled to the transmitter output.
With an input signal amplitude fed to the speech
/
WAWA
Ap7/ON
11II
min
,_/,
-MI
/i! 11
/I
100
200
300
11
RL
-
*
11111
11111
700 1000
500
2000
R,_.
3000
11
11
5000
FREQUENCY (CPS)
Figure 24
BASE ATTENUATION
CHART
Frequency attenuation caused by various
values of coupling capacitor with a grid resistor of 0.5 megohm in the following stage
(Re > RL)
amplifier of such amplitude that limiting does
not take place, a substantially clean sine wave
should be obtained on the carrier of the transmitter at all input frequencies up to the cutoff
frequency of the filter system in the speech
amplifier and of the filter which includes the
modulation transformer. Above these cutoff
frequencies very little modulation of the carrier
wave should be obtained. To obtain a check
on the effectiveness of the "built out" modulation transformer, the capacitors across the
primary and secondary should be removed for
the test. In most cases a marked deterioration
in the waveform output of the modulator will
be noticed with frequencies in the voice range
from 500 to 1500 cycles being fed into the
speech amplifier.
A filter system similar to that shown in figure 17 may be used between the modulator
and the modulated circuit in a grid -modulated
or screen -modulated transmitter. Lower -voltage
capacitors and low -current chokes may of
course be employed.
Boss Suppression
Most
of the power repre-
sented by ordinary speech
particularly the male voice) lies below 1000
cycles. If all frequencies below 400 or 500
cycles are eliminated or substantially attenuated, there is a considerable reduction in
power but insignificant reduction in intelligi(
www.americanradiohistory.com
HANDBOOK
bility. This means that the speech level may
be increased considerably without overmodulation or overload of the audio system. In
addition, if speech clipping is used, attenuation of the lower audio frequencies before the
clipper will reduce phase shift and canting of
the clipper output.
A simple method of bass suppression is to
reduce the size of the interstage coupling capacitors in a resistance coupled amplifier.
Figure 24 shows the frequency characteristics
caused by such a suppression circuit. A second simple bass suppression circuit is to
place a small a.c. - d.c. type filter choke from
grid to ground in a speech amplifier stage, as
shown in figure 25.
The systems described
Distortion
in the preceding paragraphs will have no effect
in reducing a broad signal caused by non linearity in the modulated amplifier. Even
though the modulating waveform impressed upon the modulated stage may be distortion free,
if the modulated amplifier is non -linear distortion will be generated in the amplifier. The
only way in which this type of distortion may
be corrected is by making the modulated amplifier more linear. Degenerative feedback
which includes the modulated amplifier in the
loop will help in this regard.
Plenty of grid excitation and high grid bias
will go a long way toward making a plate modulated Class C amplifier linear, although
such operating conditions will make more difficult the problem of TVI reduction. If this still
does not give adequate linearity, the preceding buffer stage may be modulated 50 per cent
or so at the same time and in the same phase
as the final amplifier. The use of a grid leak
to obtain the majority of the bias for a Class
C stage will improve its linearity.
The linearity of a grid -bias modulated r -f
amplifier can be improved, after proper adjustments of excitation, grid bias, and antenna
coupling have been made by modulating the
stage which excites the grid -modulated amplifier. The preceding driver stage may be grid bias modulated or it may be plate modulated.
Modulation of the driver stage should be in
the same phase as that of the final modulated
amplifier.
Modulated Amplifier
15 -8
Suppression
Bass
The Bias -Shift
Heising Modulator
The simple Class A modulator is limited to
an efficiency of about 30 %, and the tube must
dissipate the full power input during periods
of quiescence. Class AB and class B audio
systems have largely taken the place of the
old Heising modulator because of this great
305
.01
Figure 25
USE OF PARALLEL INDUCTANCE
FOR BASS SUPPRESSION
waste of power. It is possible, however, to
vary the operating bias of the class A modulator in such a way as to allow class A operation only when an audio signal is applied to
the grid of the tube. During resting periods,
the bias can be shifted to a higher value,
dropping the resting plate current and plate
dissipation of the tube. When voice waveforms
having
o w average power are employed, the
efficiency of the system is comparable to the
popular class B modulator.
1
The characteristic curve for a class
A
modu-
lator is shown in figure 26. Normal bias is
used, and the operating point is placed in the
middle of the linear portion of the Eg -Ip curve.
Maximum plate input is limited by the plate
dissipation of the tube under quiescent condition. The bias -shift modulator is biased
close to plate current cut -off under no signal
condition (figure 27). Resting plate current
+IP
,
LINEAR PORTION
-I
oP Ec
CURVE
EG
P
All
All!
BIAS
RESTING BIAS VOLTAGE
P
GRID INPUT
SIGNAL
Fìgur° 26
CHARACTERISTIC GRID
VOLTAGE -PLATE CURRENT
CURVE FOR CLASS A
HEISING MODULATOR
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SIGNAL
+Ec
CUT -OFF
-
PLATE
OUTPUT
/Inplitude Modulation
306
THE
RADIO
+te
CLA
S
C
AMPLIFIER
DDI P
AUDIO
AMPLIFIER
BIAS -SNIFT
MODULATOR
^
(MAX 5/GNAL)
Bf
BIAS-SHIFT
PLATE CURRENT
EXCURSION
REGT FIER
I P
(NO SIGNAL)
+Ec
EG
FILTER
--0
BIAS -SNIFT
CONTROL
TUBE
CUT-OFF
BIAS
_
BIAS -SNIFT
EXCURSION
P
NEGATIVE
BIAS
SUPPLY
CLASS A OPERATING
BIAS LINE
QUIESCENT
BIAS LINE
Figure
27
BIAS -SHIFT MODULATOR
Figure 28
BLOCK DIAGRAM OF
BIAS -SHIFT MODULATOR
OPERATING CHARACTERISTICS
Modulator is biased close to plate
current cut -off under no signal,
B. Upon application
of audio signal, the bias of the
stage is shifted toward the class
A operating point, A. Bias -shift
voltage is obtained from audio
condition,
signal.
and plate dissipation are therefore quite low.
Upon application of an audio signal, the bias
of the stage is shifted toward the class A
operating point, preventing the negative peaks
of the applied audio voltage from cutting off
the plate current of the tube. As the audio
voltage increases, the operating bias point is
shifted to the right on figure 27 until the class
A operating point is reached at maximum ex-
citation.
The bias -shift voltage may be obtained
directly from the exciting signal by rectification, as shown in figure 28. A simple low
pass filter system is used that will pass only
the syllabic components of speech. Enough
negative bias is applied to the bias -shift modulator to cut the resting plate current to the
desired value, and the output of the bias control rectifier is polarized so as to "buck"
the fixed bias voltage. No spurious modulation frequencies are generated, since the
modulator operates class A throughout the
audio cycle.
This form of grid pulsing permits the modulator stage to work with an pverall efficiency
of greater than 50 %, comparing favorably with
the class B modulator. The expensive class B
driver and output transformers are not required,
since resistance coupling may be used in the
input circuit of the bias -shift modulator, and
a heavy -duty filter choke will serve as an impedance coupler for the modulated stage.
Series and Parallel
Control Circuits
The
bias -shift
system
make take one of several
forms. A "series" control
circuit is shown in figure 29. Resting bias is
applied to the bias -shift modulator tube through
the voltage divider R2 /R4. The bias control
tube is placed across resistor R2. Quiescent
bias for the modulator is set by adjusting R2.
As the internal resistance of the bias control
tube is varied at a syllabic rate the voltage
drop across R2 will vary in unison. The modulator bias, therefore varies at the same rate.
Excitation for the bias control tube is obtained
from the audio signal through potentiometer
RI which regulates the amplitude of the control signal. The audio signal is rectified by
the bias control rectifier, and filtered by network R3 -Cl in the grid circuit of the bias con-
trol tube.
The "parallel" control system is illustrated
in figure 30. Resting bias for the modulator is
obtained from the voltage divider R2 /R4.
Potentiometer R2 adjusts the resting bias
level, determining the static plate current of
the modulator. Resistor R3 serves as a bias
resistor for the control tube, reducing its plate
current to a low level. When an audio signal
is applied via R1 to the grid of the control
tube the internal resistance is lowered, decreasing the shunt resistance across R2. The
negative modulator bias is therefore reduced.
The bias axis of the modulator is shifted from
the cut -off region to a point on the linear
portion of the operating curve. The amount of
bias -shift is controlled by the setting of
potentiometer R1. Capacitor Cl in conjunction
with bias resistor R3 form a syllabic filter for
www.americanradiohistory.com
HANDBOOK
Heising Modulator
BIAS-SHIFT
SPEECH
AMPLIFIER
TO MODULATED
B
R-F AMPLIFIER
MODULATOR
307
B4
SPEECH
BIAS -SHIFT
MODULATOR
AMPLIFIER
R
TO MODULATED
-F AMPLIFIER
ADJUST
OPERAT/N
BIAS
DJUST
OPERA
TING
BIAS
ADJUST
RES T/NG
BIAS
BIAS CONTROL
RECTIFIER
BIAS CONTROL
TUBE
BIAS CONTROL
TUBE
NEGATIVE
FtA
MODULATOR
BIAS
NEGATIVE
MODULATOR BIAS
ADJUST REST /NG
81
S
2
Figure 29
"SERIES" CONTROL CIRCUIT
Figure 30
FOR BIAS -SHIFT MODULATOR
"PARALLEL" CONTROL
The internal resistance of the
bias control tube is varied of a
syllabic rate to change the
operating bias of the modulator
CIRCUIT FOR BIAS -SHIFT
MODULATOR
The resistance to ground of point
A in the bias network is varied
at a syllabic rate by the bias
control tube.
tube.
the control bias that is applied to the modu-
lator stage.
large value of plate dissipation is required for the bias -shift modulator tube. For
plate voltages below 1500, the 211 (VT -4C)
A
may be used, while the 304 -TL is suitable for
voltages up to 3000. As with normal class A
amplifiers, low mu tubes function best in this
circuit.
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CHAPTER SIXTEEN
Frequency Modulation and
Radioteletype Transmission
Exciter systems for FM and single sideband
transmission are basically similar in that modification of the signal in accordance with the
intelligence to be transmitted is normally accomplished at a relatively low level. Then the
intelligence- bearing signal is amplified to the
desired power level for ultimate transmission.
True, amplifiers for the two types of signals
are basically different; linear amplifiers of the
Class, A or Class B type being used for ssb
signals, while Class C or non -linear Class B
amplifiers may be used for FM amplification.
But the principle of low -level generation and
subsequent amplification is standard for both
types of transmission.
16 -1
the advantages of FM for certain types of communication pointed out. Since the distinguishing features of the two types of transmission
lie entirely in the modulating circuits at the
transmitter and in the detector and limiter circuits in the receiver, these parts of the communication system will receive the major portion of attention.
Modulation is the process of altering a radio wave in accordance
with the intelligence to be transmitted. The
nature of the intelligence is of little importance as far as the process of modulation is
concerned; it is the method by which this intelligence is made to give a distinguishing
characteristic to the radio wave which will
enable the receiver to convert it back into intelligence that determines the type of modulation being used.
Figure 1 is a drawing of an r -f carrier amplitude modulated by a sine -wave audio voltage. After modulation the resultant modulated
r-f wave is seen still to vary about the zero
axis at a constant rate, but the strength of the
individual r -f cycles is proportional to the amplitude of the modulation voltage.
In figure 2, the carrier of figure 1 is shown
frequency modulated by the same modulating
voltage. Here it may be seen that modulation
voltage of one polarity causes the carrier frequency to decrease, as shown by the fact that
the individual r -f cycles of the carrier are
spaced farther apart. A modulating voltage of
the opposite polarity causes the frequency to
Modulation
Frequency Modulation
The use of frequency modulation and the
allied system of phase modulation has become
of increasing importance in recent years. For
amateur communication frequency and phase
modulation offer important advantages in the
reduction of broadcast and TV interference
and in the elimination of the costly high -level
modulation equipment most commonly employed
with amplitude modulation. For broadcast work
FM offers an improvement in signal -to-noise
ratio for the high field intensities available
in the local -coverage area of FM and TV broad-
cast stations.
In this chapter various points of difference
between FM and amplitude modulation transmission and reception will be discussed and
308
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Frequency Modulation
309
UNMODULATED CARRIER AMPLITUDE
f\,
CARRIER
SIDO FREQUENCY t SIDE FREQUENCY
FREQUENCY
Figure
J
FIGURE
FIGURE 2
AM AND FM WAVES
Figure 1 shows a sketch of the scope pattern of
an amplitude modulated wave at the bottom. The
center sketch shows the modulating wave and the
upper sketch shows the carrier wave.
Figure 2 shows at the bottom a sketch of a frequency modulated wave. In this case the center
sketch also shows the modulating wave and the
Laper sketch shows the carrier wave. Note that
the carrier wave and the modulating wave are the
same in either case, but that the waveform of the
modulated wave is quite different in the two
cases.
increase, and this is shown by the r-f cycles
being squeezed together to allow more of them
to be completed in a given time interval.
Figures 1 and 2 reveal two very important
characteristics about amplitude- and frequency- modulated waves. First, it is seen that
while the amplitude (power) of the signal is
varied in AM transmission, no such variation
takes place in FM. In many cases this advantage of FM is probably of equal or greater importance than the widely publicized noise reduction capabilities of the system. When 100
per cent amplitude modulation is obtained, the
average power output of the transmitter must
be increased by 50 per cent. This additional
output must be supplied either by the modulator itself, in the high -level system, or by
operating one or more of the transmitter stages
at such a low output level that they are capable of producing the additional output without
distortion, in the low -level system. On the
other hand, a frequency -modulated transmitter
requires an insignificant amount of power from
the modulator and needs no provision for increased power output on modulation peaks.
All of the stages between the oscillator and
the antenna may be operated as high- efficiency
Class B or Class C amplifiers or frequency
multipliers.
3
AM SIDE FREQUENCIES
For each AM modulating frequency, a pair of side
frequencies is produced. The side frequencies are
spaced away from the carrier by an amount equal
to the modulation frequency, and their amplitude
is directly proportional to the amplitude of the
modulation. The amplitude of the carrier does not
change under modulation.
The second characteristic of FM
and AM waves revealed by figures 1 and 2 is that both types
of modulation result in distortion of the r -f
carrier. That is, after modulation, the r -f cycles are no longer sine waves, as they would
be if no frequencies other than the fundamental carrier frequency were present. It may be
shown in the amplitude modulation case illustrated, that there are only two additional frequencies present, and these are the familiar
side /requencies, one located on each side of
the carrier, and each spaced from the carrier
by a frequency interval equal to the modulation frequency. in regard to frequency and amplitude, the situation is as shown in figure 3.
The strength of the carrier itself does not vary
during modulation, but the strength of the side
frequencies depends upon the percentage of
modulation. At 100 per cent modulation the
power in the side frequencies is equal to half
that of the carrier.
Under frequency modulation, the carrier wave
again becomes distorted, as shown in figure
2. But, in this case, many more than two additional frequencies are formed. The first two of
these frequencies are spaced from the carrier
by the modulation frequency, and the additional
side frequencies are located out on each side
of the carrier and are also spaced from each
other by an amount equal to the modulation frequency. Theoretically, there are an infinite
number of side frequencies formed, but, fortunately, the strength of those beyond the frequency swing of the transmitter under modulation is relatively low.
One set of side frequencies that might be
formed by frequency modulation is shown in
figure 4. Unlike amplitude modulation, the
Carrier-Wove
Distortion
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310
FM
Transmission
UNMODULATED CARRIER AMPLITUDE
CARRIER
SIDE
FREQUENCIES
IIli
ill
FREQUENCIES
Ill
FREQUENCY
Figure
4
FM SIDE FREQUENCIES
With FM each modulation frequency component
causes a large number of side frequencies to be
produced. The side frequencies are separated
from each other and the carrier by an amount equal
to the modulation frequency, but their anplitude
varies greatly as the amount of modulation is
changed. The carrier strength also varies greatly
with frequency modulation. The side frequencies
shown represent a case where the deviation each
side of the "carrier" frequency is equal to five
times the modulating frequency. Other amounts of
deviation with the same modulation frequency
would cause the relative strengths of the various
sidebands to change widely.
strength of the component at the carrier frequency varies widely in FM and it may even
disappear entirely under certain conditions.
The variation of strength of the carrier component is useful in measuring the amount of
frequency modulation, and will be discussed
in detail later in
this chapter.
One of the great advantages of FM over AM
is the reduction in noise at the receiver which
the system allows. If the receiver is made responsive only to changes in frequency, a considerable increase in signal -to -noise ratio is
made possible through the use of FM, when
the signal is of greater strength than the noise.
The noise reducing capabilities of FM arise
from the inability of noise to cause appreciable
frequency modulation of the noise -plus -signal
voltage which is applied to the detector in the
receiver.
Unlike amplitude modulation, the
term percentage modulation means
little in FM practice, unless the receiver characteristics are specified. There are, however,
three terms, deviation, modulation index, and
deviation ratio, which convey considerable
information concerning the character of the
FM wave.
Deviation is the amount of frequency shift
each side of the unmodulated carrier frequency
which occurs when the transmitter is modulated. Deviation is ordinarily measured in kilocycles, and in a properly operating FM transFM Terms
THE
RADIO
mitter it will be directly proportional to the
amplitude of the modulating signal. When a
symmetrical modulating signal is applied to
the transmitter, equal deviation each side of
the resting frequency is obtained during each
cycle of the modulating signal, and the total
frequency range covered by the FM transmitter
is sometimes known as the suing. If, for instance, a transmitter operating on 1000 kc.
has its frequency shifted from 1000 kc. to 1010
kc., back to 1000 kc., then to 990 kc., and
again back to 1000 kc. during one cycle of the
modulating wave, the deviation would be 10
kc. and the swing 20 kc.
The modulation index of an FM signal is
the ratio of the deviation to the audio modulating frequency, when both are expressed in
the same units. Thus, in the example above
if the signal is varied from 1000 kc. to 1010
kc. to 990 kc., and back to 1000 kc. at a rate
(frequency) of 2000 times. a second, the modulation index would be 5, since the deviation
(10 kc.) is 5 times the modulating frequency
(2000 cycles, or 2 kc.).
The relative strengths of the FM carrier and
the various side frequencies depend directly
upon the modulation index, these relative
strengths varying widely as the modulation
index is varied. In the preceding example, for
instance, side frequencies occur on the high
side of 1000 kc. at 1002, 1004, 1006, 1008,
1010, 1012, etc., and on the low frequency
side at 998, 996, 994, 992, 990, 988, etc. In
proportion to the unmodulated carrier strength
(100 per cent), these side frequencies have
the following strengths, as indicated by a
modulation index of 5: 1002 and 998 -33 per
cent, 1004 and 996-5 per cent, 1006 and 99436 per cent, 1008 and 992 -39 per cent, 1010
and 990 -26 per cent, 1012 and 988 -13 per
cent. The carrier strength (1000 kc.) will be
18 per cent of its unmodulated value. Changing the amplitude of the modulating signal will
change the deviation, and thus the modulation
index will be changed, with the result that the
side frequencies, while still located in the
same places, will have different strength values
from those given above.
The deviation ratio is similar to the modulation index in that it involves the ratio between a modulating frequency and deviation.
In this case, however, the deviation in question is the peak frequency shift obtained under
full modulation, and the audio frequency to be
considered is the maximum audio frequency to
be transmitted. When the maximum audio frequency to be transmitted is 5000 cycles, for
example, a deviation ratio of 3 would call for
a peak deviation of 3 x 5000, or 15 kc. at full
modulation. The noise -suppression capabilities of FM are directly related to the deviation ratio. As the deviation ratio is increased,
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Narrow
HANDBOOK
the noise suppression becomes better if the
signal is somewhat stronger than the noise.
Where the noise approaches the signal in
strength, however, low deviation ratios allow
communication to be maintained in many cases
where high- deviation -ratio FM and conventional AM are incapable of giving service.
This assumes that a narrow -band FM receiver
is in use. For each value of r -f signal -to -noise
ratio at the receiver, there is a maximum deviation ratio which may be used, beyond which
the output audio signal -to -noise ratio de-
creases.
Up to this critical deviation ratio,
however, the noise suppression becomes progressively better as the deviation ratio is in-
creased.
For high- fidelity FM broadcasting purposes,
deviation ratio of 5 is ordinarily used, the
maximum audio frequency being 15,000 cycles,
and the peak deviation at full modulation being 75 kc. Since a swing of 150 kc. is covered
by the transmitter, it is obvious that wide band FM transmission must necessarily be
confined to the v -h -f range or higher, where
room for the signals is available.
In the case of television sound, the deviation ratio is 1.67; the maximum modulation
frequency is 15,000 cycles, and the transmitter deviation for full modulation is 25 kc.
The sound carrier frequency in a standard TV
signal is located exactly 4.5 Mc. higher than
the picture carrier frequency. In the intercarrier TV sound system, which recently has
become quite widely used, this constant difference between the picture carrier and the sound
carrier is employed within the receiver to obtain an FM sub-carrier at 4.5 Mc. This 4.5
Mc. sub-carrier then is demodulated by the FM
detector to obtain the sound signal which
accompanies the picture.
a
Narrow -band
Narrow -Band
FM
t r
an
s-
mission has become standardized for use by the mobile services such as police, fire, and taxicab communication, and also on the basis of
a temporary authorization for amateur work in
portions of each of the amateur radiotelephone
bands. A maximum deviation of 15 kc. has
been standardized for the mobile and commercial communication services, while a maximum deviation of 3 kc. is authorized for amateur NBFM communication.
the transmitter is
most of them may
mission, when a
(speech or music)
Band
FM
311
swung are so small that
be ignored. In FM transcomplex modulating wave
is used, still additional
side frequencies resulting from a beating together of the various frequency components
in the modulating wave are formed. This is a
situation that does not occur in amplitude
modulation and it might be thought that the
large number of side frequencies thus formed
might make the frequency spectrum produced
by an FM transmitter prohibitively wide. Analysis shows, however, that the additional side
frequencies are of very small amplitude, and,
instead of increasing the bandwidth, modulation by a complex wave actually reduces the
effective bandwidth of the FM wave. This is
especially true when speech modulation is
used, since most of the power in voiced sounds
is concentrated at low frequencies in the vicinity of 400 cycles.
The bandwidth required in an FM receiver
is a function of a number of factors, both theoretical and practical. Basically, the bandwidth
required is a function of the deviation ratio
and the maximum frequency of modulation,
although the practical consideration of drift
and ease of receiver tuning also must be considered. Shown in figure 5 are the frequency
spectra (carrier and sideband frequencies)
associated with the standard FM broadcast
signal, the TV sound signal, and an amateur band narrow -band FM signal with full modulation using the highest permissible modulating
frequency in each case. It will be seen that
for low deviation ratios the receiver bandwidth should be at least four times the maximum frequency deviation, but for a deviation
ratio of 5 the receiver bandwidth need be only
about 2.5 times the maximum frequency deviation.
FM Transmission
Bandwidth Required by FM
As the above discussion has
indicated, many side frequencies are set up when a radio -
frequency carrier is frequency modulated; theoretically, in fact, an infinite number of side
frequencies is formed. Fortunately, however,
the amplitudes of those side frequencies falling outside the frequency range over which
16 -2
Direct FM Circuits
Frequency modulation may be obtained either
direct method, in which the frequency
of an oscillator is changed directly by the
modulating signal, or by the indirect method
which makes use of phase modulation. Phase modulation circuits will be discussed in section 16 -3.
A successful frequency modulated transmitter must meet two requirements: (1) The
frequency deviation must be symmetrical about
a fixed frequency, for symmetrical modulation
voltage. (2) The deviation must be directly
proportional to the amplitude of the modulation, and independent of the modulation frequency. There are several methods of direct
frequency modulation which will fulfill these
by the
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312
Transmission
FM
OA FM BROADCAST
THE
AUDIO
IN
DEVIATION - 75 KC.
MOD. FREQ.-15 KC.
MOD. INDEX - 5
e
Ñ
Iy
l
IS
.11
-105 -90 -75 -60 -45 -30 -15
V
1!
I!
C2 47
R,10K
y
}
R
n
OSCILLATOR IN
1.75 MC. RANGE
6BA6
1
RADIO
100K
f
^
+15 +30 +45 +60 +75 +90 +105
-T-
00 K- C31.,
1
POT.
470
K
1
O
TV SOUND
ati
R
,r
Ei;
-4514C
©
KC.
MOO. FREQ.- 15 KC.
MOD. INDEX -1.67
-30KC. -15
Vlt
fl
+15
KC
IN
KC
4-30
KC
+45 KC.
DEVIATION - 3 KC.
MOD. EREQ.- 3 KC.
AMATEUR NBFM
MOD. INDEX
<
-6KC.
-3KC.
+
-
cillator tank is varied. By applying audio
I
I
3 KC.
CENTER
FREQUENCY
Figure 5
EFFECT OF FM MODULATION INDEX
Showing the side - frequency amplitude and distribution for the three most conrnon modulation indices used in FM work. The maximum modulating
frequency and maximum deviation are shown in
each case.
requirements. Some of these methods
described in the following paragraphs.
Reactance-Tube
Modulators
Figure 6
REACTANCE -TUBE MODULATOR
This circuit is convenient for direct frequency modulation of on oscillator in the 1.75 -Mc.
range. Capacitor C, may be only the input
capacitance of the tube, or a small trimmer
capacitor may be included to permit a variation in the sensitivity of the reactance tube.
II*
fl
.0068
+150 -200V.
REGULATED
DEVIATION- 25
will
be
practical
One of the most
ways of obtaining direct fre-
quency modulation is through
the use of a reactance -tube modulator. In this
arrangement the modulator plate- cathode circuit is connected across the oscillator tank
circuit, and made to appear as either a capacitive or inductive reactance by exciting the
modulator grid with a voltage which either
leads or lags the oscillator tank voltage by
90 degrees. The leading or lagging grid voltage causes a corresponding leading or lagging
plate current, and the plate- cathode circuit
appears as a capacitive or inductive reactance
across the oscillator tank circuit. When the
transconductance of the modulator tube is
varied, by varying one of the element voltages,
the magnitude of the reactance across the os-
mod-
ulating voltage to one of the elements, the
transconductance, and hence the frequency,
may be varied at an audio rate. When properly
designed and operated, the reactance -tube
modulator gives linear frequency modulation,
and is capable of producing large amounts of
deviation.
There are numerous possible configurations
of the reactance -tube modulator circuit. The
difference in the various arrangements lies
principally in the type of phase -shifting circuit used to give a grid voltage which is in
phase quadrature with the r -f voltage at the
modulator plate.
Figure 6 is a diagram of one of the most
popular forms of reactance -tube modulators.
The modulator tube, which is usually a pentode such as a 6BA6, 6ÁU6, or 6CL6, has its
plate coupled through a blocking capacitor,
to the "hot" side of the oscillator grid
circuit. Another blocking capacitor, C2, feeds
r.f. to the phase shifting network R -C, in the
modulator grid circuit. If the resistance of R
is made large in comparison with the reactance of C, at the oscillator frequency, the cur-
C
through the R -C, combination will be
nearly in phase with the voltage across the
tank circuit, and the voltage across C, will
lag the oscillator tank voltage by almost 90
degrees. The result of the 90- degree lagging
voltage on the modulator grid is that its plate
current lags the tank voltage by 90 degrees,
and the reactance tube appears as an inductance in shunt with the oscillator inductance,
thus raising the oscillator frequency.
The phase- shifting capacitor C, can consist
of the input capacitance of the modulator tube
and stray capacitance between grid and ground.
rent
www.americanradiohistory.com
Reactance Tube
HANDBOOK
313
in figures 6 and 7. The audio input may be
applied to the suppressor grid, rather than the
AUDIO
IN
+150 -2D0
V.
REGULATED
Figure
7
ALTERNATIVE REACTANCE -TUBE
MODULATOR
This circuit is often preferable for use in the
lower frequency range, although it may be
used at 1.75 Mc. and above if desired. In the
schematic above the reactance tube is shown
connected across the voltage- divider capacitors of a Clapp oscillator, although the modulator circuit may be used with any common
type of oscillator.
However, better control of the operating conditions of the modulator may be had through
the use of a variable capacitor as C,. Resistance R will usually have a value of between
4700 and 100,000 ohms. Either resistance or
transformer coupling may be used to feed audio
voltage to the modulator grid. When a resistance coupling is used, it is necessary to shield
the grid circuit adequately, since the high impedance grid circuit is prone to pick up stray
r -f and low frequency a -c voltage, and cause
undesired frequency modulation.
An alternative reactance modulator circuit
is shown in figure 7. The operating conditions
are generally the same, except that the r -f
excitation voltage to the grid of the reactance
tube is obtained effectively through reversing
the R and C, of figure 6. In this circuit a small
capacitance is used to couple r.f. into the grid
of the reactance tube, with a relatively small
value of resistance from grid to ground. This
circuit has the advantage that the grid of the
tube is at relatively low impedance with respect to r.f. However, the circuit normally is
not suitable for operation above a few megacycles due to the shunting capacitance within
the tube from grid to ground.
Either of the reactance -tube circuits may
be used with any of the common types of oscillators. The reactance modulator of figure
6 is shown connected to the high- impedance
point of a conventional hot -cathode Hartley
oscillator, while that of figure 7 is shown connected across the low- impedance capacitors
of a series -tuned Clapp oscillator.
There are several possible variations of the
basic reactance-tube modulator circuits shown
control grid, if desired. Another modification
is to apply the audio to a grid other than the
control grid in a mixer or pentagrid converter
tube which is used as the modulator. Generally, it will be found that the transconductance
variation per volt of control -element voltage
variation will be greatest when the control
(audio) voltage is applied to the control grid.
In cases where it is desirable to separate completely the audio and r -f circuits, however,
applying audio voltage to one of the other elements will often be found advantageous despite the somewhat lower sensitivity.
One of the simplest methods
of adjusting the phase shift to
the correct amount is to place
a pair of earphones in series with the oscillator cathode -to- ground circuit and adjust the
phase -shift network until minimum sound is
heard in the phones when frequency modulation is taking place. If an electron -coupled or
Hartley oscillator is used, this method requires that the cathode circuit of the oscillator be inductively or capacitively coupled to
the grid circuit, rather than tapped on the grid
coil. The phones should be adequately bypassed for r.f. of course.
Adjusting the
Phase
Shift
Stabilization
Due to the presence of the react-
ance-tube frequency modulator,
the stabilization of an FM oscillator in regard
to voltage changes is considerably more involved than in the case of a simple self-controlled oscillator for transmitter frequency
control. If desired, the oscillator itself may be
made perfectly stable under voltage changes,
but the presence of the frequency modulator
destroys the beneficial effect of any such
stabilization. It thus becomes desirable to
apply the stabilizing arrangement to the modulator as well as the oscillator. If the oscillator itself is stable under voltage changes, it
is only necessary to apply voltage- frequency
compensation to the modulator.
Reactance -Tube
Modulators
Two simple
reactance -t u b e
modulators that may be applied
to an existing v.f.o. are illustrated in figures 8 and 9. The circuit of figure
8 is extremely simple, yet effective. Only two
tubes are used exclusive of the voltage regulator tubes which perhaps may be already incorporated in the v.f.o. A 6AU6 serves as a
high -gain voltage amplifier stage, and a 6CL6
is used as the reactance modulator since its
high value of transconductance will permit a
large value of lagging current to be drawn
under modulation swing. The unit should be
www.americanradiohistory.com
FM
314
Transmission
RADIO
THE
6AU6
(r
6CL6
50 L U
68 ULF
4.7
GRFi D OR
RFC
CATHODE
NOTE:
2.814H
ALL RESISTORS 0.5 WATT UNLESS
OTHERWISE NOTED
ALL CAPACITORS IN 1./F UNLESS
OTHERWISE NOTED
VR
ADJUST FOR CORRECT
CURRENT
Figure 8
SIMPLE FM REACTANCE -TUBE MODULATOR
mounted in close proximity to the v.f.o. so that
the lead from the 6CL6 to the grid circuit of
the oscillator can be as short as possible. A
practical solution is to mount the reactance
modulator in a small box on the side of the
v -f -o cabinet.
By incorporating speech clipping in the reactance modulator unit, a much more effective
use is made of a given amount of deviation.
When the FM signal is received on an AM receiver by means of slope detection, the use
of speech clipping will be noticed by the greatly increased modulation level of the FM signal, and the attenuation of the center frequency
null of no modulation. In many cases, it is
difficult to tell a speech -clipped FM signal
from the usual AM signal.
A more complex FM reactance modulator incorporating a speech clipper is shown in figure 9. A 12AX7 double triode speech amplifier
provides enough gain for proper clipper action
when a high level crystal microphone is used.
A double diode 6AL5 speech clipper is used,
the clipping level being set by the potentiometer controlling the plate voltage applied to the
diode. A 6CL6 serves as the reactance modu-
lator.
12AX7
The reactance modulator may best be adjusted by listening to the signal of the v -f -o
exciter at the operating frequency and adjusting the gain and clipping controls for the best
modulation level consistent with minimum side band splatter. Minimum clipping occurs when
the Adj. Clip. potentiometer is set for mayimum
voltage on the plates of the 6AL5 clipper tube.
As with the case of all reactance modulators,
a voltage regulated plate supply is required.
Linearity Test
It is almost a necessity to run
a static test on the reactance tube frequency modulator to determine its linearity and effectiveness, since small changes
in the values of components, and in stray capacitances will almost certainly alter the modulator characteristics. A frequency- versus -control -voltage curve should be plotted to ascertain that equal increments in control voltage,
both in a positive and a negative direction,
cause equal changes in frequency. If the curve
shows that the modulator has an appreciable
amount of non -linearity, changes in bias, electrode voltages, r -f excitation, and resistance
6AL5
DJUST GAIN
6CL6
500 ULF
ri
66LÚ
CHICAGO TRANS.
4.7A
LPF -2 FILTER
TO GRID OR
CAT/IODE OF
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R FC
2.SMH
5
.01
.01
NOTE'. ALL CAPACITORS IN OF UNLESS
OTHERWISE NOTED
ALL RESISTORS 0.5 WATT UNLESS
OTHERWISE NOTED
ADJUST
100A
CLIPPING
Figure
9
FM REACTANCE MODULATOR WITH SPEECH
CLIPPER
www.americanradiohistory.com
f
ADJUST FOR
CORRECT VR CURRENT
e+
Phase
HANDBOOK
TO
MODULATOR
CONTROL ELEMENT
values may be made to obtain a straight -line
characteristic.
Figure 10 shows a method of connecting
two 4/2 -volt C batteries and a potentiometer
to plot the characteristic of the modulator. It
will be necessary to use a zero-center voltmeter to measure the grid voltage, or else reverse the voltmeter leads when changing from
positive to negative grid voltage. When a
straight -line characteristic for the modulator
is obtained by the static test method, the capacitances of the various by -pass capacitors
in the circuit must be kept small to retain this
characteristic when an audio voltage is used
to vary the frequency in place of the d -c voltage with which the characteristic was plotted.
16 -3
Phase Modulation
By means of phase modulation (PM) it is
possible to dispense with self -controlled oscillators and to obtain directly crystal -controlled FM. In the final analysis, PM is sim-
ply frequency modulation in which the deviation is directly proportional to the modulation
frequency. If an audio signal of 1000 cycles
causes a deviation of % kc., for example, a
2000 -cycle modulating signal of the same amplitude will give a deviation of 1 kc., and so
on. To produce an FM signal, it is necessary
to make the deviation independent of the modulation frequency, and proportional only to the
modulating signal. With PM this is done by
including a frequency correcting network in
the transmitter. The audio correction network
must have an attenuation that varies directly
with frequency, and this requirement is easily
met by a very simple resistance- capacity network.
The only disadvantage of PM, as compared
to direct FM such as is obtained through the
use of a reactance -tube modulator, is the fact
that very little frequency deviation is produced directly by the phase modulator. The
deviation produced by a phase modulator is
independent of the actual carrier frequency on
315
which the modulator operates, but is dependent only upon the phase deviation which is
being produced and upon the modulation frequency. Expressed as an equation:
modulating frequency
Where Fd is the frequency deviation one way
from the mean value of the carrier, and M, is
the phase deviation accompanying modulation
expressed in radians(a radian is approximately
57.3 °). Thus, to take an example, if the phase
deviation is % radian and the modulating frequency is 1000 cycles, the frequency deviation
applied to the carrier being passed through
the phase modulator will be 500 cycles.
It is easy to see that an enormous amount
of multiplication of the carrier frequency is
required in order to obtain from a phase modulator the frequency deviation of 75 kc. required
for commercial FM broadcasting. However, for
amateur and commercial narrow -band FM work
(NBFM) only a quite reasonable number of
multiplier stages are required to obtain a deviation ratio of approximately one. Actually,
phase modulation of approximately one -half
radian on the output of a crystal oscillator in
the 80 -meter band will give adequate deviation
for 29 -Mc. NBFM radiotelephony. For example;
if the crystal frequency is 3700 kc., the deviation in phase produced is t/ radian, and the
modulating frequency is 500 cycles, the deviation in the 80 -meter band will be 250 cycles.
But when the crystal frequency is multiplied
on up to 29,600 kc. the frequency deviation
will also be multiplied by 8 so that the resulting deviation on the 10 -meter band will be 2 kc.
either side of the carrier for a total swing in
carrier frequency of 4 kc. This amount of deviation is quite adequate for NBFM work.
Odd -harmonic distortion is produced when
FM is obtained by the phase- modulation method, and the amount of this distortion that can
be tolerated is the limiting factor in determining the amount of PM that can be used. Since
the aforementioned frequency- correcting network causes the lowest modulating frequency
to have the greatest amplitude, maximum phase
modulation takes place at the lowest modulating frequency, and the amount of distortion
that can be tolerated at this frequency determines the maximum deviation that can be obtained by the PM method. For high -fidelity
broadcasting, the deviation produced by PM is
limited to an amount equal to about one -third
of the lowest modulating frequency. But for
NBFM work the deviation may be as high as
0.6 of the modulating frequency before distortion becomes objectionable on voice modulation. In other terms this means that phase deviations as high as 0.6 radian may be used for
amateur and commercial NBFM transmission.
Fd =
Figure 10
REACTANCE -TUBE LINEARITY CHECKER
Modulation
www.americanradiohistory.com
MP
316
FM
REACTANCE
TUBE
Transmission
CRYSTAL
OSCILLATOR
TUBE
r --
THE
y
RADIO
When a single- frequency mod ulating voltage is used with an
FM transmitter, the relative
amplitudes of the various sidebands and the
carrier vary widely as the deviation is varied
by increasing or decreasing the amount of modulation. Since the relationship between the
amplitudes of the various sidebands and carrier to the audio modulating frequency and the
deviation is known, a simple method of measuring the deviation of a frequency modulated
transmitter is possible. In making the measurement, the result is given in the form of the
modulation index for a certain amount of audio
input. As previously described, the modulation
index is the ratio of the peak frequency deviation to the frequency of the audio modulation.
The measurement is made by applying a
sine -wave audio voltage of known frequency
to the transmitter, and increasing the modulation until the amplitude of the carrier component of the frequency modulated wave reaches
zero. The modulation index for zero carrier
may then be determined from the table below.
As may be seen from the table, the first point
of zero carrier is obtained when the modulation
Measurement
NE %T
STAGE
LOW-C
AUDIO
Figure 11
REACTANCE -TUBE MODULATION OF
CRYSTAL OSCILLATOR STAGE
A simple reactance modula cor normally used for FM
may also be used for PM by
connecting it to the plate circuit of a crystal
oscillator stage as shown in figure 11.
Phase -Modulation
Circuits
Another PM circuit, suitable for operation
on 20, 15 and 10 meters with the use of 80
meter crystals is shown in figure 12. A double
triode 12AX7 is used as a combination Pierce
crystal oscillator and phase modulator. C,
should not be thought of as a neutralizing condenser, but rather as an adjustment for the
phase of the r -f voltage acting between the
grid and plate of the 12AX7 phase modulator.
C2 acts as a phase angle and magnitude control, and both these condensers should be adjusted for maximum phase modulation capabili-
ties of the circuit. Resonance of the circuit is
established by the iron slug of coil L, -L,. A
6CL6 is used as a doubler to 7 Mc. and delivers approximately 2 watts on this band. Additional doubler stages may be added after the
6CL6 stage to reach the desired band of operation.
Still another PM circuit, which is quite widely used commercially, is shown in figure 13.
In this circuit L and C are made resonant at a
frequency which is 0.707 times the operating
frequency. Hence at the operating frequency
the inductive reactance is twice the capacitive
reactance. A cathode follower tube acts as a
variable resistance in series with the L and
C which go to make up the tank circuit. The
operating point of the cathode follower should
be chosen so that the effective resistance in
series with the tank circuit (made up of the
resistance of the cathode- follower tube in parallel with the cathode bias resistor of the cathode follower) is equal to the capacitive reactance of the tank capacitor at the operating frequency. The circuit is capable of about plus
or minus % radian deviation with tolerable distortion.
of
Deviation
index has a value of 2.405, -in other words,
when the deviation is 2.405 times the modulation frequency. For example, if a modulation
frequency of 1000 cycles is used, and the
modulation is increased until the first carrier
null is obtained, the deviation will then be
2.405 times the modulation frequency, or 2.405
kc. If the modulating frequency happened to be
2000 cycles, the deviation at the first null
would be 4.810 kc. Other carrier nulls will be
obtained when the index is 5.52, 8.654, and at
increasing values separated approximately by
rr. The following is a listing of the modulation
index at successive carrier nulls up to the
tenth:
Zero carrier
Modulation
point no.
index
1
2
3
4
5
6
7
8
9
10
2.405
5.520
8.654
11.792
14.931
18.071
21.212
24353
27.494
30.635
The only equipment required for making the
measurements is a calibrated audio oscillator
of good wave form, and a communication receiver equipped with a beat oscillator and
crystal filter. The receiver should be used
with its crystal filter set for minimum bandwidth to exclude sidebands spaced from the
carrier by the modulation frequency. The un-
www.americanradiohistory.com
HANDBOOK
FM
Reception
317
6C L6
12AX7
TO DOUBLER STAGES
250 100F
L -SET RSl E. }SPACED 71 APART ON
36E. r POWDERED IRON COR
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FORM
O/A.
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300 V.
UNLESS
OTHERWISE NOTED
ALL CAPACITORS IN 4/F UNLESS
OTHERWISE NOTED
Figure 12
REACTANCE MODULATOR FOR 10, 15 AND
modulated carrier is accurately tuned in on the
receiver with the beat oscillator operating.
Then modulation from the audio oscillator is
applied to the transmitter, and the modulation
is increased until the first carrier null is obtained. This carrier null will correspond to a
modulation index of 2.405, as previously mentioned. Successive null points will correspond
to the indices listed in the table.
A volume indicator in the transmitter audio
system may be used to measure the audio level
required for different amounts of deviation,
and the indicator thus calibrated in terms of
frequency deviation. If the measurements are
made at the fundamental frequency of the oscillator, it will be necessary to multiply the
frequency deviation by the harmonic upon which
the transmitter is operating, of course. It will
probably be most convenient to make the determination at some frequency intermediate between that of the oscillator and that at which
the transmitter is operating, and then to multiply the result by the frequency multiplication
between that frequency and the transmitter
output frequency.
16 -4
A
Reception of FM
Signals
conventional communications receiver may
be used to receive narrow -band FM transmissions, although performance will be much poorer than can be obtained with an NBFM receiver
or adapter. However, a receiver specifically
designed for FM reception must be used when
it is desired to receive high deviation FM such
20
METER OPERATION
as used by FM broadcast stations, TV sound,
and mobile communications FM.
The FM receiver must have, first of all, a
bandwidth sufficient to pass the range of frequencies generated by the FM transmitter. And
since the receiver must be a superheterodyne
if it is to have good sensitivity at the frequencies to which FM is restricted, i-f bandwidth
is an important factor in its design.
The second requirement of the FM receiver
is that it incorporate some sort of device for
converting frequency changes into amplitude
changes, in other words, a detector operating
on frequency variations rather than amplitude
variations. The third requirement, and one which
is necessary if the full noise reducing capa-
65.17
R F.
PHASE- MODULATED
OUTPUT
INPUT
fo
+B 200 V.
01
AUDIO IN
XL ABOUT
XC ABOUT
1500n
AT
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fo
f0
Figure 13
CATHODE -FOLLOWER PHASE
MODULATOR
The phase modulator illustrated above is
quite satisfactory when the stage is to be
operated on a single frequency or over a narrow range of frequencies.
www.americanradiohistory.com
318
MIXE
T
FM
I. r.
AMPLIFIER
Transmission
LIMITER
FREQUENCY
DETECTOR
(DISCRIMINATOR)
THE
RADIO
AUDIO
AMP.
OSCILLATOR
F
Figure 14
FM RECEIVER BLOCK DIAGRAM
Up to the amplitude limiter stage, the FM
receiver is similar to an AM receiver, except
for a somewhat wider i -f bandwidth. The limiter removes any amplitude modulation, and
the frequency detector following the limiter
converts frequency variations into amplitude
variations.
bilities of
the FM system of transmission are
a limiting device to eliminate amplitude variations before they reach the detector. A block diagram of the essential parts
of an FM receiver is shown in figure 14.
desired, is
The simplest device for con venting frequency variations
to amplitude variations is an
"off- tune" resonant circuit, as illustrated in
figure 15. With the carrier tuned in at point
"A," a certain amount of r -f voltage will be
developed across the tuned circuit, and, as
the frequency is varied either side of this frequency by the modulation, the r -f voltage will
increase and decrease to points "C" and "B"
in accordance with the modulation. If the voltage across the tuned circuit is applied to an
The Frequency
Detector
ordinary detector, the detector output will vary
in accordance with the modulation, the amplitude of the variation being proportional to the
deviation of the signal, and the rate being
equal to the modulation frequency. It is obvious
from figure 15 that only a small portion of the
resonance curve is usable for linear conversion
R
E Q
U E N C Y
Figure 15
SLOPE DETECTION OF FM SIGNAL
One side of the response characteristic of a
tuned circuit or of on i -f amplifier may be
used as shown to convert frequency variations of an incoming signal into amplitude
variations.
of frequency variations into amplitude variations, since the linear portion of the curve is
rather short. Any frequency variation which
exceeds the linear portion will cause distortion of the recovered audio. It is also obvious
by inspection of figure 15 that an AM receiver
used in this manner is wide open to signals
on the peak of the resonance curve and also
to signals on the other side of the resonance
curve. Further, no noise limiting action is afforded by this type of reception. This system,
therefore, is not recommended for FM reception, although widely used by amateurs for
occasional NBFA1 reception.
Travis Discriminator
Another form of frequency detector or discriminator, is shown in figure 16. In this arrangement two tuned circuits are used, one tuned
on each side of the i -f amplifier frequency,
and with their resonant frequencies spaced
slightly more than the expected transmitter
swing. Their outputs are combined in a differential rectifier so that the voltage across the
series load resistors, R, and R2, is equal to
the algebraic sum of the individual output
voltages of each rectifier. When a signal at the
At its "center" frequency t he discriminator
n
produces zero
output
voltage.
either side of
Figure 16
TRAVIS DISCRIMINATOR
This type of discriminator makes use
two
off-tuned resonant circuits coupled to of
a single primary winding. The circuit is capable
of excellent linearity, but is difficult to align.
On
this
frequency It gives
a voltage of a polarity and magnitude
which depend on the
direction and amount
FREQUENCY
of frequency shift.
Figure 17
DISCRIMINATOR VOLTAGE -FREQUENCY
CURVE
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FM
HANDBOOK
Figure
18
FOSTER-SEELEY DISCRIMINATOR
This discriminator is the most widely used
circuit since it is capable of excellent linearity and is relatively simple to align when
proper test equipment is available.
i -f mid -frequency is received, the voltages
across the load resistors are equal and opposite, and the sum voltage is zero. As the r -f
signal varies from the mid -frequency, however,
these individual voltages become unequal, and
a voltage having the polarity of the larger voltage and equal to the difference between the
two voltages appears across the series resistors, and is applied to the audio amplifier.
The relationship between frequency and discriminator output voltage is shown in figure
17. The separation of the discriminator peaks
and the linearity of the output voltage vs. frequency curve depend upon the discriminator
frequency, the Q of the tuned circuits, and the
value of the diode load resistors. As the intermediate (and discriminator) frequency is increased, the peaks must be separated further to
secure good linearity and output. Within limits,
as the diode load resistance or the Q is reduced, the linearity improves, and the separation between the peaks must be greater.
The most widely used form of
discriminator is that shown in
figure 18. This type of discriminator yields an output- voltage- versus -frequency characteristic similar to that shown in figure 19. Here, again, the output voltage is equal
to the algebraic sum of the voltages developed
across the load resistors of the two diodes,
the resistors being connected in series to
ground. However, this Foster-Seeley discriminator requires only two tuned circuits instead
of the three used in the previous discriminator.
The operation of the circuit results from the
phase relationships existing in a transformer
having a tuned secondary. In effect, as a close
examination of the circuit will reveal, the primary circuit is in series, for r.f., with each
half of the secondary to ground. When the received signal is at the resonant frequency of
the secondary, the r -f voltage across the secondary is 90 degrees out of phase with that
across the primary. Since each diode is connected across one half of the secondary windFoster -Seeley
Discriminator
Reception
319
SECONDARY VOLTAGE
Figure 19
DISCRIMINATOR VECTOR DIAGRAM
A signal at the resonant frequency of the
secondary will cause the secondary voltage
to be 90 degrees out of phase with the primary voltage, as shown at A, and the resultant voltages R and R' are equal. If the signal frequency changes, the phase relationship also changes, and the resultant voltages
are no longer equal, as shown at B. A differ ential rectifier is used to give an output voltage proportional to the difference between
R and R'.
ing and the primary winding in series, the resultant r -f voltages applied to each are equal,
and the voltages developed across each diode
load resistor are equal and of opposite polarity. Hence, the net voltage between the top of
the load resistors and ground is zero. This is
shown vectorially in figure 19A where the resultant voltages R and R which are applied
to the two diodes are shown to be equal when
the phase angle between primary and secondary voltages is 90 degrees. If, however, the
signal varies from the resonant frequency, the
90- degree phase relationship no longer exists
between primary and secondary. The result of
this effect is shown in figure 1913 where the
secondary r -f voltage is no longer 90 degrees
out of phase with respect to the primary voltage. The resultant voltages applied to the two
diodes are now no longer equal, and a d -c
voltage proportional to the difference between
the r -f voltages applied to the two diodes will
exist across the series load resistors. As the
signal frequency varies back and forth across
the resonant frequency of the discriminator,
an a -c voltage of the same frequency as the
original modulation, and proportional to the
deviation, is developed and passed on to the
audio amplifier.
Ratio
Detector
One of the more recent types of FM
detector circuits, called the ratio
detector is diagrammed in figure 20.
The input transformer can be designed so that
the parallel input voltage to the diodes can be
taken from a tap on the primary of the trans-
320
FM
.0001
Transmission
THE
RADIO
RFC
6SJ
7
TO
015C R
I
M-
INATOR
+250
A.F.
OUTPUT
Figure 20
RATIO DETECTOR CIRCUIT
The parallel voltage to the diodes in a ratio
detector may be obtained from a tap on the
primary winding of the transformer or from
o third winding. Note that one of the diodes
is reversed from the system used with the
Foster -Seeley discriminator, and that the
output circuit is completely different. The
ratio detector does not have to be preceded
by o limiter, but is more difficult to align for
distortion -free output than the conventional
discriminator.
former, or this voltage may be obtained from a
tertiary winding coupled to the primary. The
r -f choke used must have high impedance at
the intermediate frequency used in the receiver,
although this choke is not needed if the transformer has a tertiary winding.
The circuit of the ratio detector appears
very similar to that of the more conventional
discriminator arrangement. However, it will be
noted that the two diodes in the ratio detector
are poled so that their d -c output voltages add,
as contrasted to the Foster -Seeley circuit
wherein the diodes are poled so that the d-c
output voltages buck each other. At the center
frequency to which the discriminator transformer is tuned the voltage appearing at the
top of the 1- megohm potentiometer will be one half the d -c voltage appearing at the a -v-c output terminal -since the contribution of each
diode will be the same. However, as the input
frequency varies to one side or the other of
the tuned value (while remaining within the
pass band of the i -f amplifier feeding the detector) the relative contributions of the two
diodes will be different. The voltage appearing
at the top of the 1- megohm volume control will
increase for frequency deviations in one direction and will decrease for frequency deviations
in the other direction from the mean or tuned
value of the transformer. The audio output voltage is equal to the ratio of the relative contributions of the two diodes, hence the name
ratio detector.
The ratio detector offers several advantages
over the simple discriminator circuit. The circuit does not require the use of a limiter preceding the detector since the circuit is inherently insensitive to amplitude modulation on
Figure
21
LIMITER CIRCUIT
One, or sometimes two, limiter stages normally precede the discriminator so that a constant signal level will be fed to the FM detector. This procedure eliminates amplitude
variations in the signal fed to the discriminator, so that it will respond only to frequency changes.
an incoming signal. This factor alone means
that the r -f and i -f gain ahead of the detector
can be much less than the conventional discriminator for the same overall sensitivity.
Further, the circuit provides a -v -c voltage for
controlling the gain of the preceding r -f and
i -f stages. The ratio detector is, however, susceptible to variations in the amplitude of the
incoming signal as is any other detector circuit except the discriminator with a limiter
preceding it, so that a -v -c should be used on
the stages preceding the detector.
Limiters
The limiter of an FM receiver using
a
conventional discriminator serves
to remove amplitude modulation and pass on
to the discriminator a frequency modulated
signal of constant amplitude; a typical circuit
is shown in figure 21. The limiter tube is operated as an i -f stage with very low plate voltage and with grid leak bias, so that it overloads quite easily. Up to a certain point the
output of the limiter will increase with an increase in signal. Above this point, however,
the limiter becomes overloaded, and further
large increases in signal will not give any increase in output. To operate successfully, the
limiter must be supplied with a large amount
of signal, so that the amplitude of its output
will not change for rather wide variations in
amplitude of the signal. Noise, which causes
little frequency modulation but much amplitude modulation of the received signal, is virtually wiped out in the limiter.
The voltage across the grid resistor varies
with the amplitude of the received signal. For
this reason, conventional amplitude modulated
signals may be received on the FM receiver
by connecting the input of the audio amplifier
to the top of this resistor, rather than to the
discriminator output. When properly filtered.
www.americanradiohistory.com
HANDBOOK
by a simple R -C
circuit, the voltage across
the grid resistor may also be used as a -v -c
voltage for the receiver. When the limiter is
FROM
DISCRIMINATOR
One of the
220
R2100
, C'C.
K,
321
TO AUDIO GRID
ME'.
T
R'
most important
factors in the design of an
FM receiver is the frequency
swing which it is intended to handle. It will
be apparent from figure 17 that if the straight
portion of the discriminator circuit covers a
wider range of frequencies than those generated by the transmitter, the audio output will
be reduced from the maximum value of which
the receiver is capable.
In this respect, the term "modulation percentage" is more applicable to the FM receiver
than it is to the transmitter, since the modulation capability of the communication system
is limited by the receiver bandwidth and the
discriminator characteristic; full utilization of
the linear portion of the characteristic amounts,
in effect, to 100 per cent modulation. This
means that some sort of standard must be agreed
upon, for any particular type of communication,
to make it unnecessary to vary the transmitter
swing to accommodate different receivers.
Two considerations influence the receiver
bandwidth necessary for any particular type of
communication. These are the maximum audio
frequency which the system will handle, and
the deviation ratio which will be employed.
For voice communication, the maximum audio
frequency is more or less fixed at 3000 to 4000
cycles. In the matter of deviation ratio, however, the amount of noise suppression which
the FM system will provide is influenced by
the ratio chosen, since the improvement in
signal -to -noise ratio which the FM system
shows over amplitude modulation is equivalent to a constant multiplied by the deviation
ratio. This assumes that the signal is somewhat stronger than the noise at the receiver,
however, as the advantages of wideband FM
in regard to noise suppression disappear when
the signal -to -noise ratio approaches unity.
On the other hand, a low deviation ratio is
more satisfactory for strictly communication
work, where readability at low signal -to -noise
ratios is more important than additional noise
suppression when the signal is already appreciably stronger than the noise.
As mentioned previously, broadcast FM practice is to use a deviation ratio of 5. When this
ratio is applied to a voice -communication system, the total swing becomes 30 to 40 kc. With
lower deviation ratios, such as are most frequently used for voice work, the swing becomes
proportionally less, until at a deviation ratio
of 1 the swing is equal to twice the highest
audio frequency. Actually, however, the re-
f
C
operating properly, a.v.c. is neither necessary
nor desirable, however, for FM reception alone.
Receiver Design
Considerations
Adapter
NBFM
L
340 uuF
730
uuF
R'47It, C' IE00uur.
R' 22 R, C' 3400 SWF.
Figure
22
75- MICROSECOND DE- EMPHASIS
CIRCUITS
The audio signal transmitted by FM and TV
stations has received high -frequency preemphasis, so that a de- emphasis c i r c u i t
should be included between the output of the
FM detector and the input of the audio
system.
ceiver bandwidth must be slightly greater than
the expected transmitter swing, since for distortionless reception the receiver must pass
the complete band of energy generated by the
transmitter, and this band will always cover a
range somewhat wider than the transmitter
swing.
Standards in FM broadcast
and TV sound work call for
the pre- emphasis of all audio modulating frequencies above about 2000
cycles, with a rising slope such as would be
produced by a 75- microsecond RL network.
Thus the FM receiver should include a compensating de- emphasis RC network with a time
constant of 75 microseconds so that the overall frequency response from microphone to
loudspeaker will approach linearity. The use
of pre- emphasis and de- emphasis in this manner results in a considerable improvement in
the overall signal -to -noise ratio of an FM system. Appropriate values for the de- emphasis
network, for different values of circuit impedance are given in figure 22.
Pre -Emphasis
and De- Emphasis
The unit diagrammed in figure
23 is designed to provide
NBFM reception when attached
to any communication receiver having a 455 -kc.
i -f amplifier. Although NBFM can be received
on an AM receiver by tuning the receiver to
one side or the other of the incoming signal,
a tremendous improvement in signal -to -noise
ratio and in signal to amplitude ratio will be
obtained by the use of a true FM detector system.
The adapter uses two tubes. A 6AU6 is
used as a limiter, and a 6AL5 as a discriminator. The audio level is approximately 10
A NBFM 455 -kc.
Adapter Unit
Radio Teletype
Transmission
FM
322
6AL5
6AÚ6
IOOK
SODUF
loo
AUDIO
OUT
UL
100K
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I.F. IN
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220K
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V
AT 3 MA.
NOTE: ALL CAPACITORS IN /JP UNLESS OTHERWISE NOTED
ALL RESISTORS 0.5 WATT UNLESS OTHERWISE NOTED
Figure 23
NBFM ADAPTER FOR 455-KC.
I
-F SYSTEM
volts peak for the maximum deviation which
can be handled by a conventional 455 -kc. i -f
system. The unit may be tuned by placing a
high resistance d -c voltmeter across R, and
tuning the trimmers of the i-f transformer for
maximum voltage when an unmodulated signal
is injected into the i -f strip of the receiver.
The voltmeter should next be connected across
the audio output terminal of the discriminator.
The receiver is now tuned back and forth across the frequency of the incoming signal,
and the movement of the voltmeter noted. When
the receiver is exactly tuned on the signal the
voltmeter reading should be zero. When the receiver is tuned to one side of center, the voltmeter reading should increase to a maximum
value and then decrease gradually to zero as
the signal is tuned out of the passband of the
receiver. When the receiver is tuned to the
other side of the signal the voltmeter should
increase to the same maximum value but in
the opposite direction or polarity, and then
fall to zero as the signal is tuned out of the
passband. It may be necessary to make small
adjustments to C, and C, to make the voltmeter read zero when the signal is tuned in
the center of the passband.
16 -5
Radio Teletype
The teletype machine is an electric typewriter that is stimulated by d.c. pulses originated by the action of a second machine. The
pulses may be transmitted from one machine
to another by wire, or by a radio signal. When
radio transmission is used, the system is
termed radio teletype (RTTY).
The d.c. pulses that comprise the teletype
signal may be converted into three basic types
of emission suitable for radio transmission.
These are: 1- Frequency shift keying (FSK),
designated as F1 emission; 2- Make -break
keying (MBK),designated as Al emission. and;
3- Audio frequency shift keying (AFSK),
designated as F2 emission.
Frequency shift keying is obtained by varying the transmitted frequency of the radio
signal a fixed amount (usually 850 cycles)
during the keying process. The shift is accomplished in discrete intervals designated
mark and space. Both types of intervals convey
information to the teletype printer. Make -break
keying is analogous to simple c -w transmission
in that the radio carrier conveys information
by changing from an off to an on condition.
Early RTTY circuits employed MBK equipment,
which is rapidly becoming obsolete since it
is inferior to the frequency shift system.
Audio frequency shift keying employs a
steady radio carrier modulated by an audio
tone that is shifted in frequency according to
the RTTY pulses. Other forms of information
transmission may be employed by a RTTY
system which also encompass the translation
of RTTY pulses into r -f signals.
The RTTY code consists of
the 26 letters of the alphabet,
the space, the line feed, the
carriage return, the bell, the upper case shift,
and the lower case shift; making a total of 32
coded groups. Numerals, punctuation, and
symbols may be taken care of in the case shift,
since all transmitted letters are capitals.
The FSK system normally employs the higher
radio frequency as the mark, and the lower
frequency as the space. This relationship
holds true in the AFSK system also. The lower
audio frequency (mark) is normally 2125 cycles
and the higher audio tone (space) is 2975
cycles, giving a frequency difference of 850
Teletype
Coding
cycles.
The Teletype
System
simple FSK teletype system
may be added to any c -w transA
mitter. The teletype keyboard
prints the keyed letters on a tape, and at the
same time generates the electrical code group
that describes the letter. The d.c. pulses are
impressed upon a distributor unit which arranges the typing and spacing pulses in proper
sequence. The resulting series of impulses
are applied to the transmitter frequency control device, which may be a reactance modulator, actuated by a polar relay.
The received signal is hetrodyned against
a beat oscillator to provide the two audio tones
which are limited in amplitude and passed
through audio filters to separate them. Rectification of the tones permits operation of a
polar relay which can provide d.c. pulses
suitable for operation of the tele- typewriter.
www.americanradiohistory.com
CHAPTER SEVENTEEN
Sideband Transmission
While single- sideband transmission (SSB)
has attracted significant interest on amateur
frequencies only in the past few years, the principles have been recognized and put to use in
various commercial applications for many years.
Expansion of single -sideband for both commercial and amateur communication has awaited the development of economical components
possessing the required characteristics (such as
sharp cutoff filters and high stability crystals)
demanded by SSB techniques. The availability
of such components and precision test equipment now makes possible the economical testing, adjustment and use of SSB equipment on
a wider scale than before. Many of the seemingly insurmountable obstacles of past years
no longer prevent the amateur from achieving
the advantages of SSB for his class of operation.
17 -1
Commercial
Applications of
SSB
Before discussion of amateur SSB equipment,
it is helpful to review some of the commercial
applications of SSB in an effort to avoid problems that are already solved.
The first and only large scale use of SSB
has been for multiplexing additional voice circuits on long distance telephone toll wires.
Carrier systems came into wide use during the
30's, accompanied by the development of high
Q toroids and copper oxide ring modulators
of controlled characteristics.
The problem solved by the carrier system
was that of translating the 300 -3000 cycle
voice band of frequencies to a higher frequency (for example, 40.3 to 43.0 kc.) for transmission on the toll wires, and then to reverse
the translation process at the receiving terminal. It was possible in some short -haul equipment to amplitude modulate a 40 kilocycle
carrier with the voice frequencies, in which
case the resulting signal would occupy a band
of frequencies between 37 and 43 kilocycles.
Since the transmission properties of wires and
cable deteriorate rapidly with increasing frequency, most systems required the bandwidth
conservation characteristics of single -sideband
transmission. In addition, the carrier wave was
generally suppressed to reduce the power
handling capability of the repeater amplifiers
and diode modulators. A substantial body of
literature on the components and circuit techniques of SSB has been generated by the large
and continuing development effort to produce
economical carrier telephone systems.
The use of SSB for overseas radiotelephony
has been practiced for several years though
the number of such circuits has been numerically small. However, the economic value of
such circuits has been great enough to warrant elaborate station equipment. It is from
these stations that the impression has been obtained that SSB is too complicated for all but
a corps of engineers and technicians to handle.
Components such as lattice filters with 40
or more crystals have suggested astronomical
expense.
323
www.americanradiohistory.com
T H
Sideband Transmission
324
e
,10lllllllllllu IIIIIIIIIIIII1III
UPPER
LOWER
SIDEBAND
I
CARRIER
SIDEDAND
FRED.
CARRIER ENVELOPE WITH
COMBLE+ MODULATING WAVE
FREQUENCY SPECTRUM WITH
COMPLEX MODULATING WA,
Figure
REPRESENTATION OF A
CONVENTIONAL AM SIGNAL
1
More recently, SSB techniques have been
used to multiplex large numbers of voice channels on a microwave radio band using equipment principally developed for telephone carrier applications. It should be noted that all
production equipment employed in these services uses the filter method of generating the
single -sideband signal, though there is a wide
variation in the types of filters actually used.
The SSB signal is generated at a low frequency and at a low level, and then translated and linearly amplified to a high level
at the operating frequency.
Considerable development effort has been
expended on high level phasing type transmitters wherein the problems of linear amplification are exchanged for the problems of
accurately controlled phase shifts. Such equipment has featured automatic tuning circuits,
servo- driven to facilitate frequency changing,
but no transmitter of this type has been sufficiently attractive to warrant appreciable production.
17 -2
Derivation of
Single -Sideband Signals
The single -sideband method of communication is, essentially, a procedure for obtaining
more efficient use of available frequency spectrum and of available transmitter capability.
As a starting point for the discussion of single- sideband signals, let us take a conventional AM signal, such as shown in figure 1,
as representing the most common method for
transmitting complex intelligence such as
voice or music.
It will be noted in figure 1 that there are
three distinct portions to the signal: the carrier, and the upper and the lower sideband
group. These three portions always are present
in a conventional AM signal. Of all these portions the carrier is the least necessary and
the most expensive to transmit. It is an actual
E
R
A
D
I
O
fact, and it can be proved mathematically (and
physically with a highly selective receiver)
that the carrier of an AM signal remains unchanged in amplitude, whether it is being modulated or not. Of course the carrier appears
to be modulated when we observe the modulated signal on a receiving system or indicator
which passes a sufficiently wide band that
the carrier and the modulation sidebands are
viewed at the same time. This apparent change
in the amplitude of the carrier with modulation is simply the result of the sidebands beating with the carrier. However, if we receive
the signal on a highly selective receiver, and if
we modulate the carrier with a sine wave of
3000 to 5000 cycles, we will readily see that
the carrier, or either of the sidebands can be
tuned in separately; the carrier amplitude, as
observed on a signal strength meter, will remain constant, while the amplitude of the sidebands will vary in direct proportion to the
modulation percentage.
Elimination of
the Carrier and
It is obvious from the pre vious discussion that the
carrier is superfluous so far
as the transmission of intelligence is concerned. It is obviously a convenience, however, since it provides a signal
at the receiving end for the sidebands to beat
with and thus to reproduce the original modulating signal. It is equally true that the transmission of both sidebands under ordinary conditions is superfluous since identically the
same intelligence is contained in both side bands. Several systems for carrier and sideband elimination will be discussed in this
chapter.
One Sideband
Power Advantage
SSB over AM
Single sideband is a very
efficient form of voice
communication by radio.
The amount of radio frequency spectrum occupied can be no greater than the frequency
range of the audio or speech signal transmitted,
whereas other forms of radio transmission require from two to several times as much spectrum space. The r -f power in the transmitted
SSB signal is directly proportional to the power
in the original audio signal and no strong
carrier is transmitted. Except for a weak pilot
carrier present in some commercial usage,
there is no r -f output when there is no audio
input.
The power output rating of a SSB transmitter is given in terms of peak envelope
pcwer (PEP) . This may be defined as the
r -m -s power at the crest of the modulation
of
www.americanradiohistory.com
HANDBOOK
Derivation
envelope. The peak envelope power of a conventional amplitude modulated signal at 100%
modulation is four times the carrier power.
The average power input to a SSB transmitter
is therefore a very small fraction of the power
input to a conventional amplitude modulated
transmitter of the same power rating.
Single sideband is well suited for long range communications because of its spectrum
and power economy and because it is less susceptible to the effects of selective fading and
interference than amplitude modulation. The
principal advantages of SSB arise from the
elimination of the high- energy carrier and
from further reduction in sideband power permitted by the improved performance of SSB
under unfavorable propagation conditions.
In the presence of narrow band man -made
interference, the narrower bandwidth of SSB
reduces the probability of destructive interference .A statistical study of the distribution
of signals on the air versus the signal strength
shows that the probability of successful communication will be the same if the SSB power
is equal to one -half the power of one of the
two a -m sidebands. Thus SSB can give from 0
to 9 db improvement under various conditions
when the total sideband power is equal in
SSB and a -m. In general, it may be assumed
that 3 db of the possible 9 db advantage will
be realized on the average contact. In this case,
the SSB -power required for equivalent performance is equal to the power in one of the
a -m sidebands. For example, this would rate
a 100 -watt SSB and a 400 watt (carrier) a -m
transmitter as having equal performance. It
should be noted that in this comparison it is
assumed that the receiver bandwidth is just
sufficient to accept the transmitted intelligence
in each case.
To help evaluate other methods of comparison the following points should be considered.
In conventional amplitude modulation two
sidebands are transmitted, each having a peak
envelope power equal to 1/4-carrier power. For
example, a 100 -watt a -m signal will have 25watt peak envelope power in each sideband,
or a total of 50 watts. When the receiver detects this signal, the voltages of the two side bands are added in the detector. Thus the detector output voltage is equivalent to that of
a 100 -watt SSB signal. This method of comparison says that a 100 watt SSB transmitter
is just equivalent to a 100 -watt a -m transmitter. This assumption is valid only when the
receiver bandwidth used for SSB is the same
as that required for amplitude modulation
KC.
KC
AUDIO SPECTRUM
4004RC'
SSB SPECTRUM
(UPPER SIOEIAMO)
`3996
KC
325
4000 KC
SSB SPECTRUM
(LOWER S/OEBAAO
)
Figure 2
RELATIONSHIP OF AUDIO AND
SSB SPECTRUMS
The single sideband components are the same
the original
the frequency
quency of the
of the various
as
audio components except that
of each is raised by the frecarrier. The relative amplitude
components remains the same.
(e.g., 6 kilocycles) , when there is no noise or
interference other than broadband noise, and
if the a -m signal is not degraded by propagation. By using half the bandwidth for SSB
reception ( e.g., 3 kilocycles) the noise is reduced 3 db so the 100 watt SSB signal becomes equivalent to a 200 watt carrier a -m
signal. It is also possible for the a -m signal
to be degraded another 3 db on the average
due to narrow band interference and poor
propagation conditions, giving a possible 4
to 1 power advantage to the SSB signal.
It should be noted that 3 db signal -to -noise
ratio is lost when receiving only one sideband
of an a -m signal. The narrower receiving bandwidth reduces the noise by 3 db but the 6 db
advantage of coherent detection is lost, leaving
a net loss of 3 db. Poor propagation will degrade this "one sideband" reception of an a -m
signal less than double sideband reception,
however. Also under severe narrow band interference conditions (e.g., an adjacent strong
signal) the ability to reject all interference on
one side of the carrier is a great advantage.
The Nature of
SSB Signal
The nature of a single
sideband signal is easily
visualized by noting that
the SSB signal components are exactly the same
as the original audio components except that
the frequency of each is raised by the frequency of the carrier. The relative amplitude of the
various components remains the same, however. (The first statement is only true for the
upper sideband since the lower sideband frequency components are the difference between
the carrier and the original audio signal).
Figure 2A, B, and C shows how the audio
spectrum is simply moved up into the radio
spectrum to give the upper sideband. The
lower sideband is the same except inverted, as
shown in figure 2C. Either sideband may be
used. It is apparent that the carrier frequency
www.americanradiohistory.com
o
THE RADIO
Sideband Transmission
326
SINGLE TONE
Figure 3
A SINGLE SINE WAVE TONE INPUT
TO A SSB TRANSMITTER RESULTS
IN A STEADY SINGLE SINE WAVE
R -F OUTFIT (A). TWO AUDIO TONES
OF EQUAL AMPLITUDE BEAT
TOGETHER TO PRODUCE HALF -SINE
WAVES AS SHOWN IN (B).
of a SSB signal can only be changed by adding or subtracting to the original carrier frequency. This is done by heterodyning, using
converter or mixer circuits similar to those
employed in a superheterodyne receiver.
It is noted that a single sine wave tone input to a SSB transmitter results in a single
steady sine wave r -f ouput, as shown in figure
3A. Since it is difficult to measure the performance of a linear amplifier with a single
tone, it has become standard practice to use
two tones of equal amplitude for test purposes. The two radio frequencies thus pro duced beat together to give the SSB envelope
shown in figure 3B. This figure has the shape
of half sine waves, and from one null to the
next represents one full cycle of the difference
frequency. How this envelope is generated is
shown more fully in figures 4A and 4B. f,
and f2 represent the two tone signals. When
a vector representing the lower frequency tone
signal is used as a reference, the other vector
rotates around it as shown, and this action
Figure 5
TWO -TONE SSB
ENVELOPE WHEN
ONE TONE HAS
TWICE THE
AMPLITUDE OF
THE OTHER.
Figure 6
THREE -TONE SSB
ENVELOPE WHEN
EQUAL TONES OF
EQUAL FREQUENCY
SPACINGS
ARE USED.
generates the SSB envelope When the two
vectors are exactly opposite in phase, the output is zero and this causes the null in the envelope. If one tone has twice the amplitude of
the other, the envelope shape is shown in
figure 5. Figure 6 shows the SSB envelope of
three equal tones of equal frequency spacings
and at one particular phase relationship. Figure
7A shows the SSB envelope of four equal
tones with equal frequency spacings and at
one particular phase relationship. The phase
relationships chosen are such that at some instant the vectors representing the several tones
are all in phase. Figure 7B shows a SSB envelope of a square wave. A pure square wave requires infinite bandwidth, so its SSB envelope
requires infinite amplitude. This emphasizes
the point that the SSB envelope shape is not
the same as the original audio wave shape, and
usually bears no similarity to it. This is because the percentage difference between the
radio frequencies is very small, even though
one audio tone may be several times the other
in terms of frequency. Speech clipping as used
r,
FREQUENCY
fi
©
12
OF
CRRiER
Figure 7A
FOUR TONE
SSB ENVELOPE
Figure 4
VECTOR REPRESENTATION OF
TWO -TONE SSB ENVELOPE
when equal tones
with equal frequency
spacings are used
www.americanradiohistory.com
Figure 7B
SSB
ENVELOPE
OF A SQUARE
WAVE.
Peak of wave reaches
infinite amplitude.
HANDBOOK
Derivation
in amplitude modulation is of no practical
value in SSB because the SSB r -f envelopes
are so different than the audio envelopes. A
heavily clipped wave approaches a square wave
and a square wave gives a SSB envelope with
peaks of infinite amplitude as shown in figure
7B.
Carrier Frequency
Stability Requirements
o
R
G
T
PUSH -PULL
AUDIO IN
Reception of a SSB
signal
is accomplished by simply
heterodyning the carrier down to zero frequency. (The conversion frequency used in the
last heterodyne step is often called the reinserted carrier). If the SSB signal is not heterodyned down to exactly zero frequency, each
frequency component of the detected audio
signal will be high or low by the amount of
this error. An error of 10 to 20 c.p s. for speech
signals is acceptable from an intelligibility
standpoint, but an error of the order of 50
c.p.s. seriously degrades the intelligibility. An
error of 20 c.p.s. is not acceptable for the
transmission of music, however, because the
harmonic relationship of the notes would be
destroyed. For example, the harmonics of 220
c.p.s. are 440, 660, 880, etc., but a 10 c.p s.
error gives 230, 450, 670, 890, etc., or 210,
430, 650, 870, etc., if the original error is on
the other side. This error would destroy the
original sound of the tones, and the harmony
between the tones.
Suppression of the carrier is common in amateur SSB work, so the combined frequency stabilities of all oscillators in both the transmitting and receiving equipment add together to
give the frequency error found in detection.
In order to overcome much of the frequency
stability problem, it is common commercial
practice to transmit a pilot carrier at a reduced amplitude. This is usually 20 db below
one tone of a two -tone signal, or 26 db below
the peak envelope power rating of the transmitter. This pilot carrier is filtered out from
the other signals at the receiver and either amplified and used for the reinserted carrier or
used to control the frequency of a local oscillator. By this means, the frequency drift of
the carrier is eliminated as an error in detection.
Advantage of SSB
with Selective Fading
327
On long distance corn -
munication
using a -m,
circuits
selective
fading often causes severe distortion and at
times makes the signal unintelligible. When
one sideband is weaker than the other, distor-
R
r
OUT
0
Figure
8
SHOWING TWO COMMON TYPES
OF BALANCED MODULATORS
that o balanced modulator changes
the circuit condition from single ended to
push -pull, or vice versa. Choice of circuit depends upon external circuit conditions since
both the (A) and B: arrangements can give
Notice
satisfactory generation of a double -sideband
suppressed- carrier signal.
tien results; but when the carrier becomes
weak and the sidebands are strong, the distortion is extremely severe and the signal may
sound like "monkey chatter." This is because
a carrier of at least twice the amplitude of
either sideband is necessary to demodulate the
signal properly. This can be overcome by using exalted carrier reception in which the carrier is amplified separately and then reinserted
before the signal is demodulated or detected.
This is a great help, but the reinserted carrier
must be very close to the same phase as the
original carrier. For example, if the reinserted
carrier were 90 degrees from the original
source, the a -m signal would be converted to
phase modulation and the usual a -m detector
would deliver no output.
The phase of the reinserted carrier is of no
importance in SSB reception and by using a
strong reinserted carrier, exalted carrier reception is in effect realized. Selective fading with
one sideband simply changes the amplitude
and the frequency response of the system and
very seldom causes the signal to become unintelligible. Thus the receiving techniques used
with SSB are those which inherently greatly
minimize distortion due to selective fading.
www.americanradiohistory.com
328
THE RADIO
Sideband Transmission
MO
slDE-
DAN:
22
VOLTAGE. -,)
z,
OUTPUT
SHUNT-QUAD
MODULATOR
BRIDGE
MODULATOR
CARRIER
vOLTAGE
SiDE-
7
BAND
OUTPUT
z,
RING
MODULATOR
-
DOUBLE-BALANCED
MODULATOR
GAI
ggiER
I
VOLTAGE
Figure 9
TWO TYPES OF DIODE BALANCED
MODULATOR
balanced modulator circuits are commonly used in carrier telephone work and in
single-sideband systems where the carrier
frequency and modulating frequency are relatively close together. Vacuum diodes, copper oxide rectifiers, or crystal diodes may be
used in the circuits.
Such
17 -3
Carrier Elimination
Circuits
Various circuits may be employed to eliminate the carrier to provide a double sideband
signal. A selective filter may follow the carrier
elimination circuit to produce a single sideband signal.
Two modulated amplifiers may be connected
with the carrier inputs 180° out of phase, and
with the carrier outputs in parallel. The car-
rier will be balanced out of the output circuit,
leaving only the two sidebands. Such a circuit is called a balanced modulator.
Any non -linear element will produce modulation. That is, if two signals are put in, sum
and difference frequencies as well as the original frequencies appear in the output. This
phenomenon is objectionable in amplifiers and
desirable in modulators or mixers.
In addition to the sum and difference frequencies, other outputs (such as twice one
frequency plus the other) may appear. All
combinations of all harmonics of each input
frequency may appear, but in general these are
of decreasing amplitude with increasing order
of harmonic. These outputs are usually rejected by selective circuits following the modulator. All modulators are not alike in the
magnitude of these higher order outputs. Balanced diode rings operating in the square law
region are fairly good and pentagrid converters
much poorer. Excessive carrier level in tube
mixers will increase the relative magnitude
of the higher order outputs. Two types of
triode balanced modulators are shown in figure
8, and two types of diode modulators in figure
9. Balanced modulators employing vacuum
tubes may be made to work very easily to a
point. Circuits may be devised wherein both
input signals may be applied to a high impedance grid, simplifying isolation and loading problems. The most important difficulties
with these vacuum tube modulator circuits
are: (1) Balance is not independent of signal
level. (2) Balance drifts with time and environment. (3) The carrier level for low "high order output" is critical, and (4) Such circuits
have limited dynamic range.
A number of typical circuits are shown in
figure 10. Of the group the most satisfactory
performance is to be had from plate modulated
triodes.
IDO
o-.1
0.I
O T
R.r
CARIRR
IN
PULL
AUDIO IN
PU SII
PLATE MODULATED BALANCED
TRIODE MODULATOR
BALANCED TRIODE MODULATOR
WITH SINGLE ENDED INPUT CIRCUITS
Figure 10
BALANCED MODULATORS
www.americanradiohistory.com
BALANCED PENTAGRID CONVERTER MODULATOR
HANDBOOK
MODULA T.
VOLTAGE
Carrier Elimination
SIDEBAND
OurPUr
CARRIER VOLTAGE
HIGH Z
MODULATING
VOLTAGE
HIGH Z
SIDEBAND
OUTPUT
LOW Z
LOW Z
MODULATING
VOLTAGE
CARRIER VOLTAGE
DOUBLE- BALANCED RING MODULATOR
329
SIDEBAND
OUTPUT
CARRIER VOLTAGE
SHUN.' QUAD MODAL ATOR
SERIES -QUAD MODULATOR
Figure 11
DIODE RING MODULATORS
Modulation in telephone carrier equipment has been very
successfully accomplished with
copper -oxide double balanced ring modulators.
More recently, germanium diodes have been
applied to similar circuits. The basic diode
ring circuits are shown in figure 11. The most
widely applied is the double balanced ring
(A) . Both carrier and input are balanced with
respect to the output, which is advantageous
when the output frequency is not sufficiently
different from the inputs to allow ready separation by filters. It should be noted that the
carrier must pass through the balanced input
and output transformers. Care must be taken
in adapting this circuit to minimize the carrier
power that will be lost in these elements. The
shunt and series quad circuits are usable when
the output frequencies are entirely different
(i.e.: audio and r.f.) The shunt quad (B)
is used with high source and load impedances
and the series quad (C) with low source and
load impedances. These two circuits may be
adapted to use only two diodes, substituting a
balanced transformer for one side of the
bridge, as shown in figure 12. It should be
noted that these circuits present a half -wave
load to the carrier source. In applying any of
these circuits, r -f chokes and capacitors must
be employed to control the path of signal and
carrier currents. In the shunt pair, for example,
a blocking capacitor is used to prevent the r -f
load from shorting the audio input.
To a first approximation, the source and
load impedances should be an arithmetical
mean of the forward and back resistances of
the diodes employed. A workable rule of
thumb is that the source and load impedances
be ten to twenty times the forward resistance
for semi -conductor rings. The high frequency
limit of operation in the case of junction and
copper-oxide diodes may be appreciably extended by the use of very low source and load
impedances.
Copper -oxide diodes suitable for carrier
Diode Ring
Modulators
.
work are normally manufactured to order. They
offer no particular advantage to the amateur,
though their excellent long -term stability is
important in commercial applications. Rectifier types intended to be used as meter rectifiers are not likely to have the balance or high
frequency response desirable in amateur SSB
transmitters.
Vacuum diodes such as the 6AL5 may be
used as modulators. Balancing the heater cathode capacity is a major difficulty except
when the 6AL5 is used at low source and load
impedance levels. In addition, contact potentials of the order of a few tenths of a volt may
also disturb low level applications (figure 13).
The double diode circuits appear attractive,
but in general it is more difficult to balance a
transformer at carrier frequency than an additional pair of diodes. Balancing potentiometers
may be employed, but the actual cause of the
unbalance is far more subtile, and cannot be
adequately corrected with a single adjustment.
A signal produced by any of the above circuits may be classified as a double sideband,
suppressed- carrier signal.
MODULATING
VOLTAGE
I_
SIDEBAND
OUTPUT
SHUNT -PAIR
MODULATOR
CARRIER VOLTAGE
O
SERIES -PAIR
MODULATOR
MODULATI NC
VOLTAGE
ARRIER VOLTAGE
Figure 12
DOUBLE -DIODE PAIRED MODULATORS
www.americanradiohistory.com
THE RADIO
Sideband Transmission
330
B
i
PUSH-PULL R
CARRIER IN
F
R.F OUT
O
OA SERIES- BALANCED DIODE MODULATOR
USING 6AL5 TUBE
6AL5
-6
-5
-4
-3
-2
-1
0
KILOCYCLES DEVIATION
R.F.CAR
RIES IN
Figure 15
BANDPASS CHARACTERISTIC OF
BURNELL S -15000 SINGLE
SIDEBAND FILTER
R.F. OUT
oRFC
AUDIO
IN
001
OB RING -DIODE MODULATOR USING 6AL5 TUBE
Figure 13
VACUUM DIODE MODULATOR CIRCUITS
Generation of
Single -Sideband Signals
17 -4
In general, there are two commonly used
methods by which a single -sideband signal may
be generated. These systems are: (I) The Filter Method, and (2) The Phasing Method.
The systems may be used singly or in combination, and either method, in theory, may
be used at the operating frequency of the
transmitter or at some other frequency with
the signal at the operating frequency being obtained through the use of frequency changers
(mixers)
.
The Filter
Method
The filter method for obtaining
signal is the classic method which has been in use by the
telephone companies for many years both for
a SSB
100-10000
SPE ECM
AMPLIFIER
I7-SO
ZOO-50001.
'1.
SPEECH
FILTER
50-53
47-SO KC.
NC
47 -50 KC
SIDEBAND
FILTER
BALANCED
MODULATOR
land -line and radio communications. The mode
of operation of the filter method is diagrammed in figure 14, in terms of components and
filters which normally would be available to
the amateur or experimenter. The output of
the speech amplifier passes through a conventional speech filter to limit the frequency
range of the speech to about 200 to 3000
cycles. This signal then is fed to a balanced
modulator along with a 50,000 -cycle first carrier from a self-excited oscillator. A low -frequency balanced modulator of this type most
conveniently may be made up of four diodes
of the vacuum or crystal type cross connected
in a balanced bridge or ring modulator circuit.
Such a modulator passes only the sideband
components resulting from the sum and difference between the two signals being fed to
the balanced modulator. The audio signal and
the 50 -kc. carrier signal from the oscillator
both cancel out in the balanced modulator so
that a band of frequencies between 47 and 50
kc. and another band of frequencies between
50 and 53 kc. appear in the output.
The signals from the first balanced modulator are then fed through the most critical
47-5014C
PHASE
INVERTER
KC
SO KC.
OSCILLATOR
BLOCK
BALANCED
MODULATOR
1750-1950
KC.
OSCILLATOR
Figure 14
DIAGRAM OF FILTER EXCITER EMPLOYING A 50 -K.C.
SIDEBAND FILTER
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HIGH -O TUNEDCIRCUIT
FOR OUTPUT IN
1100 - 2000 AC. SAND
HANDBOOK
Generation of S.S.B.
6AL5
2 12AU7
6AU6
SNUNTDIODE MODULATOR
15MH
250LU1F
01
( SO
i
331
6C4
12AU7
RF AMPLIFIER
AMASE-INVERTER
.01
33111f LO
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G
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MIC
SO KCR.
FILTE
B
002--
{
0.35
r
005
I.
+350
V
SOMA
-PULL R.F. TO BALANCED
MODULATOR FOR CONVERSION
TO 160 METERS
PUSH
12AU7
NOTE
OSCILLATOR
SO KC
:
UNLESS OTHERWISE SPECIFIED,
RESISTORS ARE 0.5 WATT.
CAPACITORS /AI 4/F.
00A
100
Figure 16
OPERATIONAL CIRCUIT FOR SSB EXCITER USING THE BURNELL
50 -KC. SIDEBAND FILTER
component in the whole system -the first sideband filter. It is the function of this first sideband filter to separate the desired 47 to 50 kc.
sideband from the unneeded and undesired 50
to 53 kc. sideband. Hence this filter must have
low attenuation in the region between 47 and
50 kc., a very rapid slope in the vicinity of
50 kc., and a very high attenuation to the
sideband components falling between 50 and
53 kilocycles.
Burnell & Co., Inc., of Yonkers, New York
produce such a filter, designated as Burnell
S -1 5,000. The passband of this filter is shown
in figure 15.
Appearing, then, at the output of the filter
is a single sideband of 47 kc. to 50 kc. This
sideband may be passed through a phase inverter to obtain a balanced output, and then
fed to a balanced mixer. A local oscillator
operating in the range of 1750 kc. to 1950 kc.
is used as the conversion oscillator. Additional
conversion stages may now be added to trans200-30001
SPEECH
AMPLIFIER
111111
200-3000
LOW 2
PHASE
INVERTER
453
SHUNT -QUAD
RING
MODULATOR
late the SSB signal to the desired frequency.
Since only linear amplification may be used,
it is not possible to use frequency multiplying
stages. Any frequency changing must be done
by the beating-oscillator technique. An operational circuit of this type of SSB exciter is
shown in figure 16.
A second type of filter-exciter for SSB may
be built around the Collins Mechanical Filter.
Such an exciter is diagrammed in figure 17.
Voice frequencies in the range of 200 -3000
cycles are amplified and fed to a low impedance phase -inverter to furnish balanced audio.
This audio, together with a suitably chosen
r -f signal, is mixed in a ring modulator, made
up of small germanium diodes. Depending
upon the choice of frequency of the r -f oscillator, either the upper or lower sideband may be
applied to the input of the mechanical filter.
The carrier, to some extent, has been rejected
by the ring modulator. Additional carrier rejection is afforded by the excellent passband
-158 RC.
-
453-458 RC.
455
MECHANICAL
F
3953
RC.
K C.
ILTER
CONVERTER
FOR OUTPUT ON
3953 R
450-453 KC.
453
K.C.
OSCILLATOR
3500 K.C.
OSCILLATOR
Figure 17
BLOCK DIAGRAM OF FILTER EXCITER EMPLOYING A 455 -KC.
MECHANICAL FILTER FOR SIDEBAND SELECTION
www.americanradiohistory.com
R.F. AMPLIFIER
WITH HIGH -Q
TUNED CIRCUIT
C.
THE RADIO
Sideband Transmission
332
CMSO
CMSO
0
10
20
CMSO
CM SO
fT -EII
CHANNEL
AT-241 CHANNEL
If CRYSTAL
SO
CRYSTAL
4II.
J
NC.
.- FREQUENCY
CARRIER
465.PMC.
Figure 18
SIMPLE CRYSTAL LATTICE FILTER
< e0
70
459 480 461 462 463 464
FREQUENCY (KC.)
characteristics of the mechanical filter. For
simplicity, the mixing and filtering operation
usually takes place at a frequency of 455 kilocycles. The single -sideband signal appearing
at the output of the mechanical filter may be
translated directly to a higher operating frequency. Suitable tuned circuits must follow
the conversion stage to eliminate the signal
from the conversion oscillator.
The heart of a filter -type SSB
exciter is the sideband filter.
Conventional coils and capacitors may be
used to construct a filter based upon standard
wave filter techniques. The Q of the filter inductances must be high when compared with
the reciprocal of the fractional bandwidth. If
a bandwidth of 3 kc. is needed at a carrier frequency of 50 kc., the bandwidth expressed in
terms of the carrier frequency is 3/50 or 6%.
This is expressed in terms of fractional bandwidth as 1/16. For satisfactory operation, the
Wave Filters
o
I
10
UPPER SI MOAN
7WE
CARRIER
FREQUENCY
30
60
246
247
248
249
250
251
252
253
254
FREQUENCY (K.C.)
Figure 19
PASSBAND OF LOWER AND UPPER
SIDEBAND MECHANICAL FILTER
235
Q of the filter inductances should be 10 times
the reciprocal of this, or 160. Appropriate Q is
generally obtained from toroidal inductances,
though there is some possibility of using iron
core solenoids between 10 kc. and 20 kc. A
characteristic impedance below 1000 ohms
should be selected to prevent distributed capacity of the inductances from spoiling overall
performance. Paper capacitors intended for
bypass work may not be trusted for stability
or low loss and should not be used in filter
circuits. Care should be taken that the levels
of both accepted and rejected signals are low
enough so that saturation of the filter inductances does not occur.
The best known filter responses have been obtained
with crystal filters. Types designed for program carrier service cut -off 80 db in less than
50 cycles. More than 80 crystals are used in
this type of filter. The crystals are cut to control reactance and resistance as well as the
resonant frequency. The circuits used are based
on full lattices.
The war -surplus low frequency crystals may
be adapted to this type of filter with some
success. Experimental designs usually synthesize a selectivity curve by grouping sharp
notches at the side of the passband. Where the
width of the passband is greater than twice
the spacing of the series and parallel resonance of the crystals, special circuit techniques must be used. A typical crystal filter using
these surplus crystals, and its approximate passband is shown in figure 18.
Crystal Filters
Filters using mechanical
resonators ha v e been
studied by a number of companies and are
offered commercially by the Collins Radio Co.
They are available in a variety of bandwidths
Mechanical Filters
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HANDBOOK
Generation of S.S.B.
333
BALANCED
SPC
AMPLIFIEEEH R
SPEECH
FILTER
MODULATO
N I
AUDIO
PHASE
20Q JOOD
OR
NETWORKS
Q2
AND 15=110*
PHASE DIFFERENCE BETWEEN
O/
PHASE DIFFERENCE BETWEEN
RI AND RI
=
PO.
AMPLIFIER STAGES
DIRECTLY TO ANTENNA SYSTEM
TO POWER
SPLITTING
BALANCED
MODULATOF
N. 2
ei
92
RADIO FRED.
PHASE
SPLITTING
NETWORK
RADIO FRED.
SIGNAL AT
CARRIER FRED.
Figure 20
BLOCK DIAGRAM OF THE "PHASING" METHOD
The phasing method of obtaining a single -sideband signal is simpler thon the filter system in regard to
the number of tubes and circuits required. The system is also less expensive in regard to the components
required, but is more critical in regard to adjustments for the transmission of a pure single -sideband signal.
at center frequencies of 250 kc. and 455 kc.
The 250 kc. series is specifically intended for
sideband selection. The selectivity attained by
these filters is intermediate between good LC
filters at low center frequencies and engineered
quartz crystal filters. A passband of two 250
kc. filters is shown in figure 19. In application
of the mechanical filters some special precautions are necessary. The driving and pick -up
coils should be carefully resonated to the operating frequency. If circuit capacities are unknown, trimmer capacitors should be used
across the coils. Maladjustment of these tuned
coils will increase insertion loss and the peak to- valley ratio. On high impedance filters ( ten
to twenty thousand ohms) signals greater than
2 volts at the input should be avoided. D -c
should be blocked out of the end coils. While
the filters are rated for 5 ma. of coil current,
they are not rated for d -c plate voltage.
a number of points
of view from which the operation of the phasing system
of SSB generation may be described. We may
state that we generate two double-sideband
suppressed carrier signals, each in its own balanced modulator, that both the r -f phase and
the audio phase of the two signals differ by
90 degrees, and that the outputs of the two
balanced modulators are added with the result
that one sideband is increased in amplitude
and the other one is cancelled. This, of course,
is a true description of the action that takes
place. But it is much easier to consider the
phasing system as a method simply of adding
(or of subtracting) the desired modulation
frequency and the nominal carrier frequency.
The carrier frequency of course is not trans-
The Phasing
System
There are
mitted, as is the case with all SSB transmissions, but only the sum or the difference of the
modulation band from the nominal carrier is
transmitted ( figure 20 ) .
The phasing system has the obvious advantage that all the electrical circuits which give
rise to the single sideband can operate in a
practical transmitter at the nominal output frequency of the transmiter. That is to say that
if we desire to produce a single sideband whose
nominal carrier frequency is 3.9 Mc., the
balanced modulators are fed with a 3.9 -Mc.
signal and with the audio signal from the
phase splitters. It is not necessary to go through
several frequency conversions in order to obtain a sideband at the desired output frequency, as in the case with the filter method
of sideband generation.
Assuming that we feed a speech signal to
the balanced modulators along with the 3900 kc. carrier (3.9 Mc.) we will obtain in the output of the balanced modulators a signal which
is either the sum of the carrier signal and the
speech band, or the difference between the carrier and the speech band. Thus if our speech
signal covers the band from 200 to 3000
cycles, we will obtain in the output a band of
frequencies from 3900.2 to 3903 kc. (the sum
of the two, or the "upper" sideband), or a band
from 3897 to 3899.8 kc. (the difference between the two or the "lower" sideband) . A
further advantage of the phasing system of
sideband generation is the fact that it is a very
simple matter to select either the upper sideband or the lower sideband for transmission.
A simple double -pole double -throw reversing
switch in two of the four audio leads to the
balanced modulators is all that is required.
www.americanradiohistory.com
Sideband Transmission
334
T H
E
R
A
D
I
O
of a few milliwatts it is most common to make
the first stage in the amplifier chain a class
A amplifier, then to use one or more class B
linear amplifiers to bring the output up to the
desired level.
R.F
OUT
o
\
180690270/
FOUR-PHASE A
F.
INDUCTIVE
COUPLING
0
IBC'
ar 270
FOUR -PHASE A. F.
Figure 21
TWO CIRCUITS FOR SINGLE
SIDEBAND GENERATION BY THE
PHASING METHOD.
The circuit of (A) offers the advantages of
simplicity in the single-ended input circuits
plus a push -pull output circuit. Circuit (8) requires double -ended input circuits but allows
all the plates to be connected in parallel for
the output circuit.
High -Level
Phasing vs.
Low -Level Phasing
Balanced
Illustrated in figure 8 are
the two basic balanced
modulator circuits which
give good results with a radio frequency carrier and an audio modulating signal. Note that
one push -pull and one single ended tank circuit is required, but that the push -pull circuit
may be placed either in the plate or the grid
circuit. Also, the audio modulating voltage always is fed into the stage in push -pull, and
the tubes normally are operated Class A.
When combining two balanced modulators
to make up a double balanced modulator as
used in the generation of an SSB signal by the
phasing system, only one plate circuit is required for the two balanced modulators. However, separate grid circuits are required since
the grid circuits of the two balanced modulators operate at an r -f phase difference of 90
degrees. Shown in figure 21 are the two types
of double balanced modulator circuits used for
generation of an SSB signal. Note that the circuit of figure 21A is derived from the balanced modulator of figure 8A, and similarly
figure 21B is derived from figure 8B.
Another circuit that gives excellent performance and is very easy to adjust is shown in
figure 22. The adjustments for carrier balance
are made by adjusting the potentiometer for
voltage balance and then the small variable capacitor for exact phase balance of the balanced
carrier voltage feeding the diode modulator.
Modulator Circuits
MECHANICAL
FILTER
The plate -circuit efficiency of the four tubes
usually used to make up
the two balanced modulators of the phasing system may run as high
as 50 to 70 per cent, depending upon the operating angle of plate current flow. Hence it is
possible to operate the double balanced modulator directly into the antenna system as the
output stage of the transmitter.
The alternative arrangement is to generate
the SSB signal at a lower level and then to
amplify this signal to the level desired by
means of class A or class B r -f power amplifiers. If the SSB signal is generated at a level
VOLT
SSB
0
1
OUTPUT
R-F CARRIER
2.3 VOLTS
BALANCED
Figure 22
MODULATOR FOR
USE
WITH MECHANICAL FILTER
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HANDBOOK
335
Generation of S.S.B.
fISO
V.
3900
.024220I1
TO BAL.
MOO. I
AUDIO
SIGNAL
3900
3900
TO SAL
00l3!
9100 .2
20 K
7
B
.150
Figure 23
LOW -Q R -F PHASE -SHIFT NETWORK
The r -f phase -shift system illustrated above
is convenient in a case where it is desired to
make small changes in the operating frequency of the system without the necessity
of being precise in the adjustment of two
coupled circuits as used for r -f phase shift
in the circuit of figure 21.
Radio-Frequency
Phasing
A single -sideband genera -
tor of the phasing type
requires that the two balanced modulators be fed with r -f signals having a 90- degree phase difference. This r -f
phase difference may be obtained through the
use of two loosely coupled resonant circuits,
such as illustrated in figures 21A and 21B.
The r -f signal is coupled directly or inductively to one of the tuned circuits, and the coupling
between the two circuits is varied until, at
resonance of both circuits, the r -f voltages
developed across each circuit have the same
amplitude and a 90- degree phase difference.
The 90- degree r -f phase difference also may
be obtained through the use of a low -Q phase
shifting network, such as illustrated in figure
23; or it may be obtained through the use of
a lumped -constant quarter -wave line. The low Q phase- shifting system has proved quite practicable for use in single -sideband systems,
particularly on the lower frequencies. In such
an arrangement the two resistances R have
the same value, usually in the range between
100 and a few thousand ohms. Capacitor C, in
shunt with the input capacitances of the tubes
and circuit capacitances, has a reactance at
the operating frequency equal to the value of
the resistor R. Also, inductor L has a net inductive reactance equal in value at the operating frequency to resistance R.
The inductance chosen for use at L must
take into account the cancelling effect of the
input capacitance of the tubes and the circuit
capacitance; hence the inductance should be
v.
Figure 24
DOME AUDIO -PHASE -SHIFT NETWORK
This circuit arrangement is convenient for obtaining the audio phase shift when it is desired
to use
a
minimum of circuit components and
tube elements.
variable and should have a lower value of inductance than that value of inductance which
would have the same reactance as resistor R.
Inductor L may be considered as being made
up of two values of inductance in parallel; (a)
a value of inductance which will resonate at
the operating frequency with the circuit and
tube capacitances, and ( b) the value of inductance which is equal in reactance to the resistance R. In a network such as shown in figure
23, equal and opposite 45- degree phase shifts
are provided by the RL and RC circuits, thus
providing a 90- degree phase difference between the excitation voltages applied to the
two balanced modulators.
Audio -Frequency
The audio -frequency phaseshifting networks used in
generating a single -sideband signal by the phasing method usually are
based on those described by Dome in an article in the December, 1946, Electronics. A
relatively simple network for accomplishing
the 90- degree phase shift over the range from
160 to 3500 cycles is illustrated in figure 24.
The values of resistance and capacitance must
be carefully checked to insure minimum deviation from a 90- degree phase shift over the 200
to 3000 cycle range.
Another version of the Dome network is
shown in figure 25. This network employs
Phasing
three 12AU7 tubes and provides balanced output for the two balanced modulators. As with
the previous network, values of the resistances
within the network must be held to very close
tolerances. It is necessary to restrict the speech
range to 300 to 3000 cycles with this network.
Audio frequencies outside this range will not
have the necessary phase -shift at the output
www.americanradiohistory.com
336
THE RADIO
Sideband Transmission
12AÚ7
12AU7
12AU7
+105 V. REGULATED
TO BAL.
MOD Y
0.5
I
R
.01
PUSH -PULL
AUDIO
INPUT
AUD
INPUT
N
K
0.5
M0D2
B
2450
VLF
Ill
607
Figure 26
11M
PASSIVE AUDIO -PHASE-SHIFT
NETWORK, USEFUL OVER RANGE
OF 300 TO 3000 CYCLES.
30
1%
+105 V. REGULATED
Figure 25
A VERSION OF THE DOME
AUDIO -PHASE-SHIFT
NETWORK
of the network and will show up as spurious
emissions on the sideband signal, and also
in the region of the rejected sideband. A low pass 3500 cycle speech filter, such as the
Chicago Transformer Co. LPF -2 should be
used ahead of this phase -shift network.
A passive audio phase -shift network that
employs no tubes is shown in figure 26. This
network has the same type of operating restrictions as those described above. Additional
information concerning phase -shift networks
will be found in Single Sideband Techniques
published by the Cowan Publishing Corp.,
New York, and The Single Sideband Digest
published by the American Radio Relay
League. A comprehensive sideband review is
contained in the December, 1956 issue of
Proceedings of the I.R.E.
Comparison of Filter
and Phasing Methods
of SSB Generation
1!s
MODSI
TO SAL.
343
2
TO BAL
MOD II 2
1)]]B
TO SAL.
Either the filter or the
phasing method of
single -sideband generation is theoretically
capable of a high degree of performance.
In general, it may be said that a high degree
of unwanted signal rejection may be attained
with less expense and circuit complexity with
the filter method. The selective circuits for
rejection of unwanted frequencies operate at a
relativly low frequency, are designed for this
one frequency and have a relatively high order
of Q. Carrier rejection of the order of 50 db or
so may be obtained with a relatively simple
filter and a balanced modulator, and unwanted
sideband rejection in the region of 60 db is
economically possible.
The phasing method of SSB generation exchanges the problems of high -Q circuits and
linear amplification for the problems of accurately controlled phase -shift networks. If the
phasing method is employed on the actual
transmitting frequency, change of frequency
must be accompanied by a corresponding rebalance of the phasing networks. In addition,
it is difficult to obtain a phase balance with
ordinary equipment within 2% over a band of
audio frequencies. This means that carrier
suppression is limited to a maximum of 40 db
or so. However, when a relatively simple SSB
transmitter is needed for spot frequency operation, a phasing unit will perform in a satisfactory manner.
Where a high degree of performance in the
SSB exciter is desired, the filter method
and the phasing method may be combined.
Through the use of the phasing method in the
first balanced modulator those undesired sideband components lying within 1000 cycles of
the carrier may be given a much higher degree
of rejection than is attainable with the filter
method alone, with any reasonable amount of
complexity in the sideband filter. Then the
sideband filter may be used in its normal way
to attain very high attenuation of all undesired
sideband components lying perhaps further
than 500 cycles away from the carrier, and to
restrict the sideband width on the desired side
of the carrier to the specified frequency limit.
Single Sideband
Frequency Conversion Systems
17 -5
In many instances the band of sideband
frequencies generated by a low level SSB transmitter must be heterodyned up to the desired
carrier frequency. In receivers the circuits
which perform this function are called converters or mixers. In sideband work they are
usually termed mixers or modulators.
Mixer Stages
One circuit which can be used
for this purpose employs a
receiving -type mixer tube, such as the 6BE6.
The output signal from the SSB generator is
fed into the #1 grid and the conversion fre-
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HANDBOOK
Frequency Conversion
337
68E6
2000 KC.
CONVERSION
FREQUENCY
(S.S
)
100
TUNE TO SELECT
2000 + 250 =2250 KC.
ON
2000 -220 1750 KC.
250 KC. SSB
SIGNAL
(0.25V. )
SSB OUTPUT
12AU7
2.0 VOLT
CONVERSION
0.2 VOLT
SIGNAL INPUT
SIGNAL
Figure 27
PENTAGRID MIXER CIRCUIT FOR
SSB FREQUENCY CONVERSION
Figure 28
TWIN TRIODE MIXER CIRCUIT FOR
quency into the #3 grid. This is the reverse of
the usual grid connections, but it offers about
10 db improvement in distortion. The plate
circuit is tuned to select the desired output
frequency product. Actually, the output of the
mixer tube contains all harmonics of the two
input signals and all possible combinations of
the sum and difference frequencies of all the
harmonics. In order to avoid distortion of the
SSB signal, it is fed to the mixer at a low
level, such as 0.1 to 0.2 volts. The conversion
frequency is fed in at a level about 20 db
higher, or about 2 volts. By this means, harmonics of the incoming SSB signal generated
in the mixer tube will be very low. Usually
the desired output frequency is either the sum
or the difference of the SSB generator carrier
frequency and the conversion frequency. For
example, using a SSB generator carrier frequency of 250 kc. and a conversion injection
frequency of 2000 kc. as shown in figure 27,
the output may be tuned to select either 2250
kc. or 1750 kc.
Not only is it necessary to select the desired
mixing product in the mixer output but also
the undesired products must be highly attenuated to avoid having spurious output signals
from the transmitter. In general, all spurious
signals that appear within the assigned frequency channel should be at least 60 db below
the desired signal, and those appearing outside of the assigned frequency channel at least
80 db below the signal level.
When mixing 250 kc. with 2000 kc. as in
the above example, the desired product is the
2250 kc. signal, but the 2000 kc. injection
frequency will appear in the output about 20
db stronger than the desired signal. To reduce
it to a level 80 db below the desired signal
means that it must be attenuated 100 db.
The principal advantage of using balanced
modulator mixer stages is that the injection
frequency theoretically does not appear in the
output. In practice, when a considerable frequency range must be tuned by the balanced
modulator and it is not practical to trim the
SSB
FREQUENCY CONVERSION
push -pull circuits and the tubes into exact
amplitude and phase balance, about 20 db
of injection frequency cancellation is all that
can be depended upon. With suitable trimming adjustments the cancellation can be made
as high as 40 db, however, in fixed frequency
circuit s.
The Twin Triode Mixer
The mixer circuit
shown in figure 28
has about 10 db lower distortion than the conventional 6BE6 converter tube. It has a lower
voltage gain of about unity and a lower output impedance which loads the first tuned
circuit and reduces its selectivity. In some applications the lower gain is of no consequence
but the lower distortion level is important
enough to warrant its use in high performance
equipment. The signal -to- distortion ratio of
this mixer is of the order of 70 db compared
to approximately 60 db for a 6BE6 mixer
when the level of each of two tone signals is
0.5 volt. With stronger signals, the 6BE6
distortion increases very rapidly, whereas the
12AU7 distortion is much better comparatively.
6AS6's
001
SSB
SIGNAL
INPUT
-BIAS
CARRIER + 120 V.
IN
ATSMA.
Figure 29
BALANCED MODULATOR CIRCUIT
FOR SSB FREQUENCY CONVERSION
338
THE RADIO
Sideband Transmission
1
7
6
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3
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IIIII1111EMY
9
.
.
.
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.
.
.
.
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6
6
i
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64
Figure 30
RESPONSE OF "N" NUMBER OF TUNED CIRCUITS,
ASSUMING EACH CIRCUIT Q IS 50
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HANDBOOK
Frequency Conversion
In practical equipment where the injection
frequency is variable and trimming adjustments and tube selection cannot be used, it
may be easier and more economical to obtain
this extra 20 db of attenuation by using an
extra tuned circuit in the output than by using
a balanced modulator circuit. A balanced modulator circuit of interest is shown in figure
29, providing a minimum of 20 db of carrier
attenuation with no balancing adjustment.
Selective Tuned Circuits
The selectivity requirements of the
tuned circuits following a mixer stage often
become quite severe. For example, using an
input signal at 250 kc. and a conversion injection frequency of 4000 kc. the desired output may be 4250 kc. Passing the 4250 kc.
signal and the associated sidebands without
attenuation and realizing 100 db of attenuation
at 4000 kc. (which is only 250 kc. away) is
a practical example. Adding the requirement
that this selective circuit must tune from 2250
kc. to 4250 kc. further complicates the basic
requirement. The best solution is to cascade a
number of tuned circuits. Since a large number of such circuits may be required, the most
practical solution is to use permeability tuning, with the circuits tracked together. An example of such circuitry is found in the Collins
KWS -1 sideband transmitter.
If an amplifier tube is placed between each
tuned circuit, the overall response will be the
sum of one stage multiplied by the number
of stages (assuming identical tuned circuits).
Figure 30 is a chart which may be used to
determine the number of tuned circuits required for a certain degree of attenuation at
some nearby frequency. The Q of the circuits
is assumed to be 50, which is normally realized
in small permeability tuned coils. The number
of tuned circuits with a Q of 50 required for
providing 100 db of attenuation at 4000 kc.
while passing 4250 kc. may be found as fol-
lows:
of is 4250 -4000 =250 kc.
fr is the resonant frequency, 4250 kc.
and
ff =
4250
-
0.059
The point on the chart where .059 intersects 100 db is between the curves for 6 and 7
tuned circuits, so 7 tuned circuits are required.
Another point which must be considered in
practice is the tuning and tracking error of
the circuits. For example, if the circuits were
339
actually tuned to 4220 kc. instead of 4250 kc.,
the
f'f would
be 4220 or 0.0522. Checking
the curves shows that 7 circuits would just
barely provide 100 db of attenuation. This
illustrates the need for very accurate tuning
and tracking in circuits having high attenuation properties.
Coupled Tuned
When as many as 7 tuned
circuits are required for proper attenuation, it is not
necessary to have the gain that 6 isolating amplifier tubes would provide. Several vacuum
tubes can be eliminated by using two or three
coupled circuits between the amplifiers. With
a coefficient of coupling between circuits 0.5
of critical coupling, the overall response is
very nearly the same as isolated circuits. The
gain through a pair of circuits having 0.5
coupling is only eight -tenths that of two critically coupled circuits, however. If critical
coupling is used between two tuned circuits,
the nose of the response curve is broadened
and about 6 db is lost on the skirts of each
pair of critically coupled circuits. In some
cases it may be necessary to broaden the nose
of the response curve to avoid adversely affecting the frequency response of the desired
passband. Another tuned circuit may be required to make up for the loss of attenuation
Circuits
on the skirts of critically coupled circuits.
Frequency Conversion
Problems
The example in the
previous section shows
the difficult selectivity problem encountered when strong undesired
signals appear near the desired frequency. A
high frequency SSB transmitter may be required to operate at any carrier frequency in
the range of 1.75 Mc. to 30 Mc. The problem
is to find a practical and economical means of
heterodyning the generated SSB frequency to
any carrier frequency in this range. There are
many modulation products in the output of the
mixer and a frequency scheme must be found
that will not have undesired output of appreciable amplitude at or near the desired signal.
When tuning across a frequency range some
products may "cross over" the desired frequency. These undesired crossover frequencies
should be at least 60 db below the desired
signal to meet modern standards. The amplitude of the undesired products depends upon
the particular characteristics of the mixer and
the particular order of the product. In general,
most products of the 7th order and higher
will be at least 60 db down. Thus any cross-
www.americanradiohistory.com
THE RADIO
Sideband Transmission
340
FROM SSB
GENERATOR
GAIN CONTROL
PREAMPLIFIER
y
"SIGNAL TO
DISTORTION
(S /D) RATIO
3P
2V 2A-Q
ANT.
PLATE CIRCUIT
R F
RECTIFIER
t
11
SP-4Q 4P-3R
+TO
STAGE
R-F FROM P-A
CONTROL BIAS
II
POWER
AMPLIFIER
I
P
Q
2Q7 3Q4j
431.
DELAY BIAS VOLTAGE
FROM POWER SUPPLY
5t41.
Figure 31
SSB DISTORTION PRODUCTS,
SHOWN UP TO NINTH ORDER
Figure 32
BLOCK DIAGRAM OF AUTOMATIC
LOAD CONTROL (A.L.C.) SYSTEM
over frequency lower than the 7th must be
avoided since there is no way of attenuating
them if they appear within the desired pass band. The General Electric Ham News, volume
11 #6 of Nov. -Dec., 1956 covers the subject
of spurious products and incorporates a "mix selector" chart that is useful in determining
spurious products for various different mixing
loaded, these spurious frequencies can extend
far outside the original channel width and
cause an unintelligible "splatter" type of interference in adjacent channels. This is usually
of far more importance than the distortion of
the original tones with regard to intelligibility
or fidelity. To avoid interference in another
channel, these distortion products should be
down at least 40 db below adjacent channel
signal. Using a two-tone test, the distortion is
given as the ratio of the amplitude of one
test tone to the amplitude of a third order
product. This is called the signal -to- distortion
ratio (S /D) and is usually given in decibels.
The use of feedback r -f amplifiers make S/D
ratios of greater than 40 db possible and practical.
schemes.
In general, for most applications when the
intelligence bearing frequency is lower than
the conversion frequency, it is desirable that
the ratio of the two frequencies be between 5
to 1 and 10 to 1. This is a compromise between avoiding low order harmonics of this
signal input appearing in the output, and
minimizing the selectivity requirements of the
circuits following the mixer stage.
17 -6
Distortion Products
Due to Nonlinearity of
R -F
is
Amplifiers
When the SSB envelope of a voice signal
distorted, a great many new frequencies are
generated. These represent all of the possible
combinations of the sum and difference frequencies of all harmonics of the original frequencies. For purposes of test and analysis,
two equal amplitude tones are used as the
SSB audio source. Since the SSB radio frequency amplifiers use tank circuits, all distortion products are filtered out except those
which lie close to the desired frequencies.
These are all odd order products; third order,
fifth order, etc.. The third order products are
2p -q and 2q -p where p and q represent the
two SSB r -f tone frequencies. The fifth order
products are 3p -2q and 3q -2p. These and
some higher order products are shown in
figure 31. It should be noted that the frequency spacings are always equal to the difference frequency of the two original tones.
Thus when a SSB amplifier is badly over-
Two means may be used to
keep the amplitude of these
distortion products down to
acceptable levels. One is to design the amplifier for excellent linearity over its amplitude
or power range. The other is to employ a
means of limiting the amplitude of the SSB
envelope to the capabilities of the amplifier.
An automatic load control ssytem (ALC) may
be used to accomplish this result. It should be
noted that the r.f wave shapes of the SSB signal are always sine waves because the tank circuits make them so. It is the change in gain
with signal level in an amplifier that distorts
the SSB envelope and generates unwanted distortion products. An ALC system may be used
to limit the input signal to an amplifier to
prevent a change in gain level caused by excessive input level.
The ALC system is adjusted so the power
amplifier is operating near its maximum power
capability and at the same time is protected
from being over -driven. In amplitude modulated systems it is common to use speech compressors and speech clipping systems to perform this function. These methods are not
Automatic
Load Control
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HANDBOOK
Distortion Products
the signal is nearly up to the full power capability of the amplifier. At this signal level,
the rectified output overcomes the delay bias
and the gain of the preamplifier is reduced
rapidly with increasing signal so that there is
very little rise in output power above the
threshold of gain control.
When a signal peak arrives that would normally overload the power amplifier, it is desireable that the gain of the ALC amplifier be
reduced in a few milliseconds to a value where
overloading of the power amplifier is overcome. After the signal peak passes, the gain
should return to the normal value in about
one -tenth second. These attack and release
times are commonly used for voice communications. For this type of work, a dynamic range
of at least 10 db is desirable. Input peaks as
high as 20 db above the threshold of compression should not cause loss of control although some increase in distortion in the upper range of compression can be tolerated because peaks in this range are infrequent. Another limitation is that the preceding SSB
generator must be capable of passing signals
above full power output by the amount of
compression desired. Since the signal level
through the SSB generator should be maintained within a limited range, it is unlikely
that more than 12 db ALC action will be
useful. If the input signal varies more than
this, a speech compressor should be used to
limit the range of the signal fed into the SSB
generator.
Figure 33 shows the effectiveness of the
ALC in limiting the output signal to the capabilities of the power amplifier. An adjustment
of the delay bias will place the threshold of
compression at the desired power output. Figure 34 shows a simplified schematic of an ALC
system. This ALC uses two variable gain am-
DB SIGNAL LEVEL INPUT
Figure 33
PERFORMANCE CURVE OF
A.L.C. CIRCUIT
equally useful in SSB. The reason for this is
that the SSB envelope is different from the
audio envelope and the SSB peaks do not
necessarily correspond with the audio peaks
as explained earlier in this chapter. For this
reason a "compressor" of some sort located
between the SSB generator and the power amplifier is most effective because it is controlled
by SSB envelope peaks rather than audio peaks.
Such a "SSB signal compressor" and the means
of obtaining its control voltage comprises a
satisfactory ALC system.
The ALC Circuit
A block diagram of an ALC
circuit is shown in figure
32. The compressor or gain control part of
this circuit uses one or two stages of remote
cutoff tubes such as 6BA6, operating very similarly to the intermediate frequency stages of
a receiver having automatic volume control.
The grid bias voltage which controls the
gain of the tubes is obtained from a voltage
detector circuit connected to the power amplifier tube plate circuit. A large delay bias is
used so that no gain reduction takes place until
6BA6
Figure 34
SIMPLIFIED SCHEMATIC OF AUTOMATIC
LOAD CONTROL AMPLIFIER. OPERATING
POINT OF ALC
CIRCUIT MAY BE
SET BY
6BA6
ee
INPUT
VARYING
BLOCKING BIAS ON
CATHODE OF 6X4
SIGNAL RECTIFIER
SENS.
ALC
341
=COMPRESSION
INDICATOR
ZERO
ADJ.
_
www.americanradiohistory.com
342
TH
Sideband Transmission
R
E
A
D
I
O
and through r -f filter capacitors. The 3.3K
resistor and 0.1 µfd. capacitor across the rectifier output stabilizes the gain around the ALC
loop to prevent "motor- boating."
17 -7
SSB JR.
R -F and A -F
Figure 35
MODULATOR CIRCUIT
sources are applied
to balanced modulator.
in series
plifier stages and the maximum overall gain is
about 20 db. A meter is incorporated which is
calibrated in db of compression. This is useful in adjusting the gain for the desired
amount of load control. A capacity voltage
divider is used to step down the r -f voltage
at the plate of the amplifier tube to about 50
volts for the ALC rectifier. The output of the
ALC rectifier passes through R -C networks
to obtain the desired attack and release times
Sideband Exciters
Some of the most popular sideband exciters
in use today are variations of the simple phasing circuit introduced in the November, 1950
issue of General Electric Ham Netes. Called the
SSB, Jr., this simple exciter is the basis for
many of the phasing transmitters now in use.
Employing only three tubes, the SSB, Jr. is a
classic example of sideband generation reduced to its simplest form.
This phasing exciter employs
audio and r -f phasing circuits
to produce a SSB signal at one spot frequency.
The circuit of one of the balanced modulator
stages is shown in figure 35. The audio signal
and r -f source are applied in series to two germanium diodes serving as balanced modulators
The SSB, Jr.
XTAL
PHASE SHIFT NETWORK
r
T2
BLV
FEO-WN/TE
voF
O.
i
12AU7
AUDIO
INPUT
000
RFC
250
,,
0.5 MH
f 101b
12ÁU7
I
2A17
TWIST
6AG7
-10.5V.
C+,6-
6.3
V.
G1, 2,3,4= INS GERMANIUM DIODE OR EQUIVALENT
C2A,B,C,D =EACH SECTION 20LF, 450 V. ELECTROLYTIC
C7= 2430 UUFD (.002 UFD MICA 5% WITH 170 -700 UUFD TRIMMER) LI, L2= 33 T. N21 E. WIRE CLOSEWOUND ON MILLEN N69046
IRON CORE ADJUSTABLE SLUG COIL FORM. LINK OF 6
CA =4600 UUFD. (.0043 UFD MICA 8.5% WITH 170-760 UUFD TRIMMER)
TURNS OF HOOKUP WIRE WOUND ON OPEN END.
C9= 1215 UUFD (.001 UFO MICA ±5% WITH 50 -360 UUFD TRIMMER)
5% WITH 5 -100 UUFD TRIMMER)L3 =16 T. N'19 E. WIRE SPACED TO FILL MILLEN M. 69046
C10 =607.5 UUFD (500 UUFD MICA
FORM. TAP AT 6 TURNS. LINK OF TURN AT CENTER.
COIL
PARALLEL)
AND
100UUFD
MICA
*10%
(2S0UUFD
600V.
C16= 35018UFD
L4 =SAME AS L, EXCEPT NO LINK USED.
1%
R7,RID= 133,300 OHMS, 1/2 WATT
LS = 26 T. OF N19 E. WIRE. LINK ON END TO MATCH LOAD.
1%
Re R9. 100,000 OHMS, 1/2 WATT
(4 TURN LINK MATCHES 72 OHM LOAD)
T1= STANCON A -53C TRANSFORMER.
T2,T3= UTC R -364 TRANSFORMER.
R- = MOUNTING ENO OF COILS
S1= DPDT TOGGLE SWITCH
f
t
1
t
t
Figure 36
SCHEMATIC, SSB, JR.
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HANDBOOK
S.B. Exc iters
having a push -pull output circuit tuned to the
r -f "carrier" frequency. The modulator drives
a linear amplifier directly at the output frequency. The complete circuit of the exciter
is shown in figure 36.
The first tube, a 12AU7, is a twin -triode
serving as a speech amplifier and a crystal
oscillator. The second tube is a I2AT7, acting as a twin channel audio amplifier following the phase-shift audio network. The linear
amplifier stage is a 6AG7, capable of a peak
power output of 5 watts.
Sideband switching is accomplished by the
reversal of audio polarity in one of the audio
channels (switch SI), and provision is made
for equalization of gain in the audio channels
(R,z) . This adjustment is necessary in order
to achieve normal sideband cancellation, which
may be of the order of 35 db or better. Phase shift network adjustment may be achieved by
adjusting potentiometer R5. Stable modulator
balance is achieved by the balance potentiometers R,° and
in conjunction with the
germanium diodes.
The SSB, Jr. is designed for spot frequency
operation. Note that when changing frequency
L,, L_, L.,, L, and L should be readjusted,
since these circuits constitute the tuning adjustments of the rig. The principal effect of
mistuning Ls, L., and L. will be lower output.
The principal effect of mistuning L , however,
will be degraded sideband suppression.
Power requirements of the SSB, Jr. are 300
volts at 60 ma., and -10.5 volts at 1 ma.
Under load the total plate current will rise to
about 80 ma. at full level with a single tone
input. With speech input, the total current
will rise from the resting value of 60 ma. to
about 70 ma., depending upon the voice waveform.
R
The "Ten -A"
Exciter
The Model 10 -A phasing exciter produced by Central
Electronics, Inc. is an advanced version of the SSB, Jr. incorporating
extra features such as VFO control, voice operation, and multi -band operation. A simplified schematic of the Model 10A is shown in
figure 37. The 12AX7 two stage speech amplifier excites a transformer coupled 1/2 -12BH7
low impedance driver stage and a voice operated (VOX) relay system employing a 12AX7
and a 6AL5. A transformer coupled 12AT7
follows the audio phasing network, providing
two audio channels having a 90- degree phase
difference. A simple 90- degree r -f phase shift
network in the plate circuit of the 9 Mc. crys-
343
tal oscillator stage works into the matched,
balanced modulator consisting of four 1N48
diodes.
The resulting 9 Mc. SSB signal may be converted to the desired operating frequency in a
6BA7 mixer stage. Eight volts of r-f from an
external v -f -o injected on grid #1 of the
6BA7 is sufficient for good conversion efficiency and low distortion. The plate circuit of
the 6BA7 is tuned to the sum or difference
mixing frequency and the resulting signal is
amplified in a 6AG7 linear amplifier stage.
Two "tweet" traps are incorporated in the
6BA7 stage to reduce unwanted responses of
the mixer which are apparent when the unit is
operating in the 14 Mc. band. Band -changing
is accomplished by changing coils L. and L°
and the frequency of the external mixing signal. Maximum power output is of the order
of 5 watts at any operating frequency.
A Simple 80 Meter
Phasing Exciter
A SSB exciter employ -
ing r -f and audio phasing circuits is shown
in figure 38. Since the r -f phasing circuits are
balanced only at one frequency of operation,
the phasing exciter is necessarily a single frequency transmitter unless provisions are made
to re- balance the phasing circuits every time a
frequency shift is made. However for mobile
operation, or spot frequency operation a relatively simple phasing exciter may be made to
perform in a satisfactory manner.
A 12AU7 is employed as a Pierce crystal
oscillator, operating directly on the chosen
SSB frequency in the 80 meter band. The second section of this tube is used as an isolation
stage, with a tuned plate circuit, L. The output of the oscillator stage is link coupled to a
90° r -f phase -shift network wherein the audio
signal from the audio phasing network is combined with the r -f signals. Carrier balance
is accomplished by adjustment of the two 1000
ohm potentiometers in the r -f phase network.
The output of the r -f phasing network is coupled through L_ to a single 6CL6 linear amplifier which delivers a 3 watt peak SSB signal on 80 meters.
A cascade 12AT7 and a single 6C4 comprise
the speech amplifier used to drive the audio
phase shift network. A small inter -stage transformer is used to provide the necessary 180°
audio phase shift required by the network. The
output of the audio phasing network is coupled to a 12AU7 dual cathode follower which
provides the necessary low impedance circuit
to match the r -f phasing network. A double-
www.americanradiohistory.com
344
Sideband Transmission
THE RADIO
Figure 37
SIMPLIFIED SCHEMATIC OF "TEN -A" EXCITER
www.americanradiohistory.com
HANDBOOK
S.B. Exciters
so*
Boo K
12 AU)
111,8
345
6CL6
L3
L2
3
f100
K
3W PEAK
SSB
150
Tuur
C
L1,
L2,L38 24T0/22E.0w
NOTE. UNLESS OTHERWISE SPECIFIED/
XR-30 PORN
.001
(0.3. O/A.)
CO/LS ARE
T EACH.
.001
ALL RESISTORS 0.3 WATT
CARRIER
EN-D4-1N71
INJECT.
1.0
70K
2511
SIDE BAN
ALL CAPACITORS /N//P.
ASTERISK AFTER CAPACITOR OR
RESISTOR VALUE IND/CATES
PRECIS /ON UNIT EAACT VALUE
CRITICAL ONLY /N T/IAT /T SHOULD
MATCH THE MAT/NG UNIT CLOSELY.
D
SELECTOR
SWITCH
200
500K
K
12AT7
STANCOR
6C 4
1
MIC.
2
12AÚ7
A53-C
-5
14
7
M
10.2, r.
SIMPLE
3 -WATT
Figure 38
PHASING TYPE
pole double -throw switch in the output circuit
of the cathode follower permits sideband selection.
A Filter -Type Exciter
for 80 and 40 Meters
A simple SSB filter -
type exciter employing the Collins mechanical filter illustrates many of the basic
principles of sideband generation. Such an exciter is shown in figure 39. The exciter is designed for operation in the 80 or 40 meter
phone bands and delivers sufficient output to
drive a class ABI tetrode such as the 2E26, 807,
or 6146. A conversion crystal may be employed, or a separate conversion v -f -o can be
used as indicated on the schematic illustration.
The exciter employs five tubes, exclusive
of power supply. They are: 6U8 low frequency
oscillator and r -f phase inverter, 6BA6 i -f amplifier, 6BA7 high frequency mixer, 6AG7
linear amplifier, and 12AU7 speech amplifier
and cathode follower. The heart of the exciter
is the balanced modulator employing two
1N81 germanium diodes and the 455 kc.,
3500 -cycle bandwidth mechanical filter. The
input and output circuits of the filter are resonated to 455 kc. by means of small padding
capacitors.
A series -tuned Clapp oscillator covers the
range of 452 kc.
457 kc. permitting the
-
+300 V.
e
500
SSB
EXCITER
carrier frequency to be adjusted to the "20
decibel" points on the response curve of the
filter, as shown in figure 40. Proper r -f signal balance to the diode modulator may be
obtained by adjustment of the padding capacitor in the cathode circuit of the triode section
of the 6U8 r -f tube. Carrier balance is set by
means of a 50K potentiometer placed across
the balanced modulator.
One half of a 12AU7 serves as a speech amplifier delivering sufficient output from a
high level crystal microphone to drive the second half of the tube as a low impedance cathode follower, which is coupled to the balanced
modulator. The two 1N81 diodes act as an
electronic switch, impressing a double sideband, suppressed- carrier signal upon the mechanical filter. By the proper choice of frequency of the beating oscillator, the unwanted
sideband may be made to fall outside the pass band of the mechanical filter. Thus a single
sideband suppressed- carrier signal appears at
the output of the filter. The 455 kc. SSB signal is amplified by a 6BA6 pentode stage, and
is then converted to a frequency in the 80
meter or 40 meter band by a 6BA7 mixer
stage. Either a crystal or an external v -f -o may
be used for the mixing signal.
To reduce spurious signals, a double tuned
www.americanradiohistory.com
B.r
w-EV
,
&MIN =WM
p
e
WM
ME
OWE
'o
I
MOW
I
,D
452
453
454
455
456
457
456
FREQUENCY (KC.)
Figure 40
THE "TWENTY DB" CARRIER
POINTS ON THE FILTER CURVE
N
Ç
III
(
1,11
>
,
The beating oscillator
p
I
IF
O
passbond. The carrier of the SSB signal is
thus attenuated 20 db in addition to the
OVN.
inherent carrier attenuation of the balanced
mixer. A total carrier attenuation of 50 db
is achieved. Unwanted sideband rejection is
of the some order.
LL
3
In
should be adjusted so
that its frequency corresponds to the 20 db
attenuation points of the mechanical filter
"
6BE 6
R
FC
500 RC.
CARRIER
INJECTION
AUDIO
SIGNAL
556
SIGNAL
FROM
F AMP.
Figure 41
THE PRODUCT DETECTOR
The above configuration resembles pentagrid
converter circuit.
®
ICYCLEOF -
MWAV
FORM
ODULATIN
ß.
LYWER
SIDEBAND
UPPER
SIDEBAND
CARRIER
FREQ.
FREQUENCY SPECTRUM WITN
COMPLEX MODULATING WAVE
Figure 39
SCHEMATIC, FILTER -TYPE SSB EXCITER
FOR 80 OR 40 METER OPERATION
DOUBLE SIDE -BAND OUTPUT
FROM BALANCED MODULATOR
WITH SINE-WAVE MODULATION
Figure 42
DOUBLE -SIDEBAND
SUPPRESSED- CARRIER SIGNAL
The envelope shown at B also is obtained on
when two audio frequencies
oscilloscope
the
of the some amplitude are fed to the input
of a single -sideband transmitter.
www.americanradiohistory.com
S.B.
transformer is placed between the mixer stage
and the 6AG7 output stage. A maximum signal of 3 watts may be obtained from the 6AG7
linear amplifier.
Selection of the upper or lower sideband is
accomplished by tuning the 6U8 beating oscillator across the passband of the mechanical
filter, as shown in figure 40. If the 80 meter
conversion oscillator is placed on the low frequency side of the SSB signal, placing the
6U8 beating oscillator on the low frequency
side of the passband of the mechanical filter
will produce the upper sideband on 80 meters.
When the beating oscillator is placed on the
high frequency side of the passband of the
mechanical filter the lower sideband will be
generated on 80 meters. If the 80 meter conversion oscillator is placed on the high frequency side of the SSB signal, the sidebands
will be reversed from the above. The variable
oscillator should be set at approximately the
20 db suppression point of the passband of
the mechanical filter for best operation, as
shown in figure 40. If the oscillator is closer
in frequency to the filter passband than this,
carrier rejection will suffer. If the oscillator is
moved farther away in frequency from the
passband, the lower voice frequencies will be
attenuated, and the SSB signal will sound high pitched and tinny. A little practice in setting
the frequency of the beating oscillator while
monitoring the 80 meter SSB signal in the
station receiver will quickly acquaint the operator with the proper frequency setting of the
beating oscillator control for transmission of
either sideband.
If desired, an amplitude modulated signal
with full carrier and one sideband may be
transmitted by placing the 6U8 low frequency
oscillator just inside either edge of the passband of the filter (designated "AM point',
figure 40) .
After the 6U8 oscillator is operating over
the proper frequency range it should be possible to tune the beating oscillator tuning capacitor across the passband of the mechanical filter and obtain a reading on the S -meter of a
receiver tuned to the filter frequency and coupled to the input grid of the 6BA6 i -f amplifier tube. The two carrier balance controls of
the 6U8 phase inverter section should be adjusted for a null reading of the S-meter when
the oscillator is placed in the center of the
filter passband. The 6BA6 stage is now
checked for operation, and transformed T1
aligned to the carrier frequency. It may be
necessary to unbalance temporarily potentio-
Exciters
347
meter #2 of the 6U8 phase inverter in order
to obtain a sufficiently strong signal for proper alignment of Ti.
A conversion crystal is next plugged in the
6BA7 conversion oscillator circuit, and the operation of the oscillator is checked by monitoring the crystal frequency with a nearby receiver. The SSB "carrier" produced by the unbalance of potentiometer #2 should be heard
at the proper sideband frequency in either the
80 meter or 40 meter band. The coupled circuit between the 6BA7 and the 6AG7 is resonated for maximum carrier voltage at the
grid of the amplifier stage. Care should be
taken that this circuit is tuned to the sideband
frequency and not to the frequency of the conversion oscillator. Finally, the 6AG7 stage is
tuned for maximum output. When these adjustments have been completed, the 455 kc.
beating oscillator should be moved just out
of the passband of the mechanical filter. The
80 meter "carrier" will disappear. If it does
not, there is either energy leaking around the
filter, or the amplifier stages are oscillating.
Careful attention to shielding (and neutralization) should cure this difficulty.
Audio excitation is now applied to the exciter, and the S-meter of the receiver should
kick up with speech, but the audio output of
the receiver should be unintelligible. As the
frequency of the beating oscillator is adjusted
so as to bring the oscillator frequency within
the passband of the mechanical filter the modulation should become intelligible. A single
sideband a.m. signal is now being generated.
The BFO of the receiver should now be turned
on, and the beating oscillator of the exciter
moved out of the filter passband. When the
receiver is correctly tuned, clean, crisp speech
should be heard. The oscillator should be set
at one of the "20 decibel" points of the filter
curve, as shown in figure 40 and all adjustments trimmed for maximum carrier suppression.
17 -8
Reception of
Single Sideband Signals
Single-sideband signals may be received,
after a certain degree of practice in the technique, in a quite adequate and satisfactory
manner with a good communications receiver.
However, the receiver must have quite good
frequency stability both in the high- frequency
oscillator and in the beat oscillator. For this
reason, receivers which use a crystal -controlled first oscillator are likely to offer a
www.americanradiohistory.com
348
TH
Sideband Transmission
greater degree of satisfaction than the more
common type which uses a self -controlled oscillator.
Beat oscillator stability in most receivers
is usually quite adequate, but many receivers
do not have a sufficient amplitude of beat oscillator injection to allow reception of strong
SSB signals without distortion. In such receivers it is necessary either to increase the
amount of beat- oscillator injection into the
diode detector, or the manual gain control of
the receiver must be turned down quite low.
The tuning procedure for SSB signals is as
follows: The SSB signals may first be located
by tuning over the band with receiver set
for the reception of c -w.; that is, with the manual gain at a moderate level and with the beat
oscillator operating. By tuning in this manner
SSB signals may be located when they are far
below the amplitude of conventional AM signals on the frequency band. Then after a signal has been located, the beat oscillator should
be turned off and the receiver put on a.v.c.
Following this the receiver should be tuned
for maximum swing of the S meter with modulation of the SSB signal. It will not be possible
to understand the SSB signal at this time, but
the receiver may be tuned for maximum deflection. Then the receiver is put back on manual
gain control, the beat oscillator is turned on
again, the manual gain is turned down until
the background noise level is quite low, and
the beat oscillator control is varied until the
signal sounds natural.
The procedure in the preceding paragraph
may sound involved, but actually all the steps
except the last one can be done in a moment.
However, the last step is the one which will
require some practice. In the first place, it is
not known in advance whether the upper or
lower sideband is being transmitted. So it will
be best to start tuning the beat oscillator from
one side of the pass band of the receiver to
the other, rather than starting with the beat
oscillator near the center of the pass band as
is normal for c -w reception.
With the beat oscillator on the wrong side
of the sideband, the speech will sound inverted;
that is to say that low- frequency modulation
tones will have a high pitch and high- frequency modulation tones will have a low pitch
and the speech will be quite unintelligible.
With the beat oscillator on the correct side of
the sideband but too far from the correct position, the speech will have some intelligibility
but the voice will sound quite high pitched.
Then as the correct setting for the beat oscilla-
-
E
R
A
D
I
O
tor is approached the voice will begin to sound
natural but will have a background growl on
each syllable. At the correct frequency for the
beat oscillator the speech will clear completely and the voice will have a clean, crisp quality. It should also be mentioned that there is
a narrow region of tuning of the beat oscillator
a small distance on the wrong side of the side band where the voice will sound quite bassy
and difficult to understand.
With a little experience it will be possible
to identify the sound associated with improper
settings of the beat -oscillator control so that
corrections in the setting of the control can be
made. Note that the main tuning control of the
receiver is not changed after the sideband
once is tuned into the pass band of the receiver. All the fine tuning should be done with
the beat oscillator control. Also, it is very important that the r -f gain control be turned to
quite a low level during the tuning process.
Then after the signal has been tuned properly
the r -f gain may be increased for good signal
level, or until the point is reached where best
oscillator injection becomes insufficient and
the signal begins to distort.
Greatly simplified tuning,
coupled with strong attenuation of undesired signals, can be obtained
through the use of a single -sideband receiver
or receiver adapter. The exalted carrier principle usually is employed in such receivers, with
a phase -sensitive system sometimes included
for locking the local oscillator to the frequency
of the carrier of the incoming signal. In order
for the locking system to operate, some carrier
must be transmitted along with the SSB signal.
Such receivers and adapters include a means
for selecting the upper or lower sideband by
the simple operation of a switch. For the reception of a single -sideband signal the switch
obviously must be placed in the correct position. But for the reception of a conventional
AM or phase- modulated signal, either sideband
may be selected, allowing the sideband with
the least interference to be used.
Single -Sideband
Receivers and
Adopters
An unusually satisfactory form of demodulator for SSB service is the product detector, shown in one form in figure 41. This
circuit is preferred since it reduces intermodulation products and does not require a large
local carrier voltage, as contrasted to the more
common diode envelope detector. This product
detector operates much in the same manner as
The Product Detector
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HANDBOOK
S.S.B. Reception
4 -250A
S4WS850
TURRET
.001
ANT.
10 N V.
200
349
00=
605
MODULATORS
250
1500
IS KV.
STANCOR
A-782
O01
10Kv
(!/SE PRIMARY
AS SECONDARY
200
-
h_
4-250A
ES 4000 V.
Figure 43
HIGH -LEVEL DSB BALANCED MODULATOR
a multi -grid mixer tube. The SSB signal is
applied to the control grid of the tube and
the locally generated carrier is impressed upon
the other control grid. The desired audio output signal is recovered across the plate resistance of the demodulator tube. Since the cathode current of the tube is controlled by the
simultaneous action of the two grids, the current will contain frequencies equal to the sum
and difference between the sideband signal and
the carrier. Other frequencies are suppressed by
the low -pass r -f filter in the plate circuit of the
stage, while the audio frequency is recovered
from the i -f sideband signal.
17 -9
Double Sideband
Transmission
Many systems of intelligence transmission
lie in the region between amplitude modulation on the one hand and single sideband suppressed- carrier transmission on the other hand.
One system of interest to the amateur is the
Synchronous Communications System, popularly known as "double sideband" (DSB -) transmission, wherein a suppressed -carrier double
sideband signal is transmitted (figure 42) .
Reception of such a signal is possible by utilizing a local oscillator phase -control system
which derives carrier phase information from
the sidebands alone and does not require the
use of any pilot carrier.
The DSB Transmitter
8 may be
Synchronous
Detection
A DSB signal may be received with difficulty on a
conventional receiver, a n d
one of the two sidebands may easily be received
on a single sideband receiver. For best reception, however, a phase- locked local oscillator
and a synchronous detector should be employed. This operation may be performed either at the frequency of reception or at a convenient intermediate frequency. A block diaI
-SYN-
Low
I-AUDIO
FILTER
AMPLIFIER
I
CHRONOUS
DETECTOR
PASS
E- DS 8
SIGNAL
LOCAL
"OSCILLATOR
FREQUENCY
CONTROL
AUDIO
0- DISCR
PHASE
IMIN.
AUDIO
AMPLIFIER
V0
PHASE
SHIFTER
A balanced modulator
of the type shown in
employed to create a DSB
signal. For higher operating levels, a pair of
class -C type tetrode amplifier tubes may be
screen modulated by a push -pull audio system
figure
and excited from a push -pull r -f source. The
plates of such a modulator are connected in
parallel to the tank circuit, as shown in figure
43. This DSB modulator is capable of 1 -kilowatt peak power output at a plate potential of
4000 volts. The circuit is self- neutralizing and
the tune -up process is much the same as with
any other class -C amplifier stage. As in the
case of SSB, the DSB signal may also be generated at a low level and amplified in linear
stages following the modulator.
0. -SYN-
CHRONOUS
DETECTOR
Q -Low
PASS
FILTER
Q -AUDIO
AMPLIFIER
Figure 44
BLOCK DIAGRAM OF DSB
RECEIVING ADAPTER
www.americanradiohistory.com
350
Sideband Transmission
gram of a DSB synchronous receiver is shown
in figure 44. The DSB signal is applied to
two detectors having their local oscillator conversion voltages in phase quadrature to each
other so that the audio contributions of the
upper and lower sidebands reinforce one another. The in -phase oscillator voltage is adjusted to have the same phase as the suppressed
carrier of the transmitted signal. The I- amplifier audio output, therefore, will contain the
demodulated audio signal, while the Q- amplifier (supplied with quadrature oscillator voltage) will produce no output due to the quadrature null. Any frequency change of the local
oscillator will produce some audio output in
the Q-amplifier, while the I- amplifier is relatively unaffected. The Q- amplifier audio will
have the same polarity as the I- channel audio
for one direction of oscillator drift, and opposite polarity for oscillator drift in the opposite
direction. The Q- amplifier signal level is proportional to the magnitude of the local oscillator phase angle error (the oscillator drift) for
small errors. By combining the I- signal and
the Q- signal in the audio phase discriminator
a d -c control voltage is developed which automatically corrects for local oscillator phase errors. The reactance tube therefore locks the
local oscillator to the correct phase. Phase control information is derived entirely from the
sideband component of the signal and the
carrier (if present) is not employed. Phase
control ceases with no modulation of the signal and is reestablished with the reappearance
of modulation.
The Beam
Deflection Modulator
17 -10
A recent development in the single side band field is the beam deflection tube (type
7360). This miniature tube employs a simple
electron "gun" which generates, controls, and
accelerates a beam of electrons directed toward
identical plates. The total plate current is determined by the voltages applied to the control
grid and screen grid of the "gun ". The division of plate current between the two plates
is determined by the difference in voltage between two deflecting electrodes placed between
the "gun" and the plates. R.f. voltage is used
to modulate the control grid of the electron
"gun" and the electron stream within the tube
may be switched between the plates by means
of an audio signal applied to the deflecting
electrodes. The 7360 makes an excellent
balanced modulator (figure 45) or product
detector having high impedance input
circuits, low distortion, and excellent carrier
suppression.
7360
.00,
I.
SO
Interference Rejection
Interference falling
within the passband
of the receiver can be reduced by proper combination of the I- and Q- audio signals. Under
phase lock conditions, the I- signal is composed of the audio signal plus the undesired
interference, whereas the Q- signal contains
only the interference component. Phase cancellation obtained by combining the two signals will reduce the interference while still
adding the desired information contained in
both side -bands. The degree of interference rejection is dependent upon the ratio of interference falling upon the two sidebands of the
received signal and upon the basic design of
the audio networks. A schematic and description of a complete DSB receiving adapter is
shown in the June, 1957 issue of CQ magazine.
R.F
IN
470 K
- PUSH-PULL
AUDIO
1.265
V.
Figure 45.
BALANCED MODULATOR CIRCUIT
USING 7360 BEAM DEFLECTION TUBE.
www.americanradiohistory.com
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CHAPTER EIGHTEEN
Transmitter Design
The excellence of a transmitter is a function of the design, and is dependent upon the
execution of the design and the proper choice
of components. This chapter deals with the
study of transmitter circuitry and of the basic
components that go to make up this circuitry.
Modern components are far from faultless. Resistors have inductance and distributed capacity. Capacitors have inductance and resistance, and inductors have resistance and
distributed capacity. None of these residual
attributes show up on circuit diagrams, yet
they are as much responsible for the success
or failure of the transmitter as are the necessary and vital bits of resistance, capacitance
and inductance. Because of these unwanted
attributes, the job of translating a circuit on
paper into a working piece of equipment often
becomes an impossible task to those individuals who disregard such important trivia.
Rarely do circuit diagrams show such pitfalls
as ground loops and residual inductive coupling between stages. Parasitic resonant circuits are rarely visible from a study of the
schematic. Too many times radio equipment is
rushed into service before it has been entirely
checked. The immediate and only too apparent
results of this enthusiasm are transmitter instability, difficulty of neutralization, r.f. wandering all over the equipment, and a general
"touchiness" of adjustment. Hand in glove
with these problems go the more serious ones
of TVI, key -clicks, and parasitics. By paying
attention to detail, with a good working knowledge of the limitations of the components,
and with a basic conception of the actions of
ground currents, the average amateur will be
able to build equipment that will work "just
like the book says."
The twin problems of TVI and parasitics
are an outgrowth of the major problem of overall circuit design. If close attention is paid
to the cardinal points of circuitry design, the
secondary problems of TVI and parasitics
will in themselves be solved.
18-1
Resistors
The resistance of a conductor is a function
of the material, the form the material takes,
the temperature of operation, and the frequency of the current passing through the resistance. In general, the variation in resistance
due to temperature is directly proportional to
the temperature change. With most wire -wound
resistors, the resistance increases with temperature and returns to its original value when
the temperature drops to normal. So- called
composition or carbon resistors have less
reliable temperature /resistance characteristics. They usually have a positive temperature coefficient, but the retrace curve as the
resistor is cooled is often erratic, and in
352
www.americanradiohistory.com
Resistors
+3
353
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10
30
40
50
SO
70
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b -20
100
DEGREES CENTIGRADE
many cases the resistance does not return to
heat cycle. It is for
this reason that care must be taken when soldering composition resistors in circuits that
require close control of the resistance value.
Matched resistors used in phase- inverter service can be heated out of tolerance by the act
of soldering them into the circuit. Long leads
should be left on the resistors and a long nose pliers should grip the lead between the
iron and the body of the resistor to act as a
heat block. General temperature characteristics of typical carbon resistors are shown
in figure 1. The behavior of an individual re-
Figure
Roc
C-
10
20
30
40
50
!O
70
SO
SO
100
HEAT CYCLE OF CONDITIONED
COMPOSITION RESISTORS
sistor will vary from these curves depending
upon the manufacturer, the size and wattage
of the resistor, etc.
Inductance of
Resistors
Every resistor because of its
physical size has in addition
to its desired resistance, less
desirable amounts of inductance and distributed capacitance. These quantities are illustrated in figure 2A, the general equivalent circuit of a resistor. This circuit represents the
actual impedance network of a resistor at any
frequency. At a certain specified frequency
2
a
L
-Ms--1001
0
1
HEAT CYCLE OF UNCONDITIONED
COMPOSITION RESISTORS
a
O
DEGREES CENTIGRADE
Figure
its original value after
-
6-O
6
_"`
,--`
`,__-_-`
``IZEIII--
` _
__-=__-_"mg_
EQUIVALENT CIRCUIT OF
A
A RESISTOR
3
M.
2
0
S
IO
FREQUENCY (MC.)
EQUIVALENT CIRCUIT OF A RESISTOR
AT A PARTICULAR FREQUENCY
Figure 3
FREQUENCY EFFECTS ON SAMPLE
COMPOSITION RESISTORS
IS
354
THE
Transmitter Design
.
WEI! -""
i-1
R=55000n
so
s
xo
I
=MINIM
20
i0
5
FREQUENCY (MC.)
Figure
4
CURVES OF THE IMPEDANCE OF WIRE WOUND RESISTORS AT RADIO
FREQUENCIES
the impedance of the resistor may be thought
of as a series reactance (X,) as shown in figure 2B. This reactance may be either inductive or capacitive depending upon whether the
residual inductance or the distributed capacitance of the resistor is the dominating factor.
As a rule, skin effect tends to increase the
reactance with frequency, while the capacity
between turns of a wire -wound resistor, or capacity between the granules of a composition
resistor tends to cause the reactance and resistance to drop with frequency. The behavior
of various types of composition resistors over
a large frequency range is shown in figure 3.
By proper component design, non -inductive resistors having a minimum of residual reactance characteristics may be constructed. Even
these have reactive effects that cannot be ignored at high frequencies.
Wirewound resistors act as low -Q inductors
at radio frequencies. Figure 4 shows typical
curves of the high frequency characteristics
of cylindrical wirewound resistors. In addition
to resistance variations wirewound resistors
exhibit both capacitive and inductive reactance, depending upon the type of resistor and
the operating frequency. In fact, such resistors perform in a fashion as low -Q r -f chokes
below their parallel self -resonant frequency.
18-2
Capacitors
The inherent residual characteristics of capacitors include series resistance, series inductance and shunt resistance, as shown in
figure 5. The series resistance and inductance
RsHUN'
M;
o--C
L
RADIO
---
RSERIES
Figure 5
EQUIVALENT CIRCUIT OF
A
CAPACITOR
depend to a large extent upon the physical
configuration of the capacitor and upon the
material of which it is made. Of great interest
to the amateur constructor is the series inductance of the capacitor. At a certain frequency the series inductive reactance of the
capacitor and the capacitive reactance are
equal and opposite, and the capacitor is in
itself series resonant at this frequency. As
the operating frequency of the circuit in which
the capacitor is used is increased above the
series resonant frequency, the effectiveness
of the capacitor as a by-passing element deteriorates until the unit is about as effective
as a block of wood.
The usual forms of by -pass capacitors have dielectrics of paper,
mica, or ceramic. For audio work,
and low frequency r -f work up to perhaps 2 Mc.
or so, the paper capacitors are satisfactory
as their relatively high internal inductance has
little effect upon the proper operation of the
circuit. The actual amount of internal inductance will vary widely with the manufacturing
process, and some types of paper capacitors
have satisfactory characteristics up to a frequency of 5 Mc. or so.
When considering the design of transmitting
equipment, it must be remembered that while
the transmitter is operating at some relatively
low frequency of, say, 7 Mc., there will be
harmonic currents flowing through the various
by -pass capacitors of the order of 10 to 20
times the operating frequency. A capacitor
that behaves properly at 7 Mc. however, may
offer considerable impedance to the flow of
these harmonic currents. For minimum harmonic generation and radiation, it is obviously
of greatest importance to employ by-pass capacitors having the lowest possible internal
By -Pass
Capacitors
inductance.
Mica dielectric capacitors have much less
internal inductance than do most paper condensers. Figure 6 lists self- resonant frequencies of various mica capacitors having various
lead lengths. It can be seen from inspection
of this table that most mica capacitors become self- resonant in the 12 -Mc. to 50 -Mc.
region. The inductive reactance they would
offer to harmonic currents of 100 Mc. or so
www.americanradiohistory.com
HANDBOOK
CONDENSER
.02
Capacitors
LEAD LENGTHS
RESONANT FREQ.
LF MICA
NONE
44.5 MC.
.002 OF MICA
NONE
23.5
MC.
10
MC.
fT
55
MC.
24
MC
55
MC.
ff
220
MC.
NO
MC.
LF MICA
.01
.000911F MICA
.002 LF CERAMIC
.001
500
LF CERAMIC
1111F
BUTTON
.001
LF CERAMIC
.01
1JF
CERAMIC
ff
NONE
14.5 MC.
Figure 6
SELF -RESONANT FREQUENCIES OF
VARIOUS CAPACITORS WITH
RANDOM LEAD LENGTH
would be of considerable magnitude. In certain
instances it is possible to deliberately series resonate a mica capacitor to a certain frequency somewhat below its normal self -resonant
frequency by trimming the leads to a critical
length. This is sometimes done for maximum
by- passing effect in the region of 40 Mc. to
60 Mc.
The recently developed button -mica capacitors shown in figure 7 are especially designed
to have extremely low internal inductance.
Certain types of button -mica capacitors of
small physical size have a self-resonant frequency in the region of 600 Mc.
Ceramic dielectric capacitors in general
have the lowest amount of series inductance
per unit of capacitance of these three universally used types of by -pass capacitors. Typical resonant frequencies of various ceramic
units are listed in figure 6. Ceramic capacitors are available in various voltage and capacity ratings and different physical configurations. Stand-off types such as shown in figure
7 are useful for by- passing socket and transformer terminals. Two of these capacitors may
be mounted in close proximity on a chassis
and connected together by an r -f choke to form
a highly effective r-f filter. The inexpensive
"clamshell" type of ceramic capacitor is
recommended for general by- passing in r-f circuitry, as it is effective as a by -pass unit to
well over 100 Mc.
The large TV "doorknob" capacitors are
useful as by -pass units for high voltage lines.
These capacitors have a value of 500 micromicrofarads, and are available in voltage ratings up to 40,000 volts. The dielectric of
these capacitors is usually titanium- dioxide.
This material exhibits piezo -electric effects,
and capacitors employing it for a dielectric
will tend to "talk- back" when a -c voltages
are applied across them. When these capaci-
355
tors are used as plate bypass units in a modulated transmitter they will cause acoustical
noise. Otherwise they are excellent for general r -f work.
A recent addition to the varied line of capacitors is the coaxial or " Hypass" type of
capacitor. These capacitors exhibit superior
by-passing qualities at frequencies up to 200
Mc. and the bulkhead type are especially effective when used to filter leads passing
through partition walls between two stages.
Variable Air
Capacitors
Even though air is the perfect
dielectric, air capacitors exhibit
losses because of the inherent
resistance of the metallic parts that make up
the capacitor. In addition, the leakage loss
across the insulating supports may become of
some consequence at high frequencies. Of
greater concern is the inductance of the capacitor at high frequencies. Since the capacitor must be of finite size, it will have tie -rods
and metallic braces and end plates, all of which
contribute to the inductance of the unit. The
actual amount of the inductance will depend
upon the physical size of the capacitor and
the methbd used to make contact to the stator
and rotor plates. This inductance may be cut
to a minimum value by using as small a capacitor as is practical, by using insulated tie rods to prevent the formation of closed inductive loops in the frame of the unit, and by
making connections to the centers of the plate
assemblies rather than to the ends as is com-
.i
y
d
Figure 7
TYPES OF CERAMIC AND MICA CAPACITORS SUITABLE FOR HIGH -FREQUENCY
BYPASSING
The Centralab 858S (1000 mad) is recoin.
mended for screen and plate circuits of tetrode tubes.
www.americanradiohistory.com
356
monly done. A large transmitting capacitor
may have an inherent inductance as large as
0.1 microhenry, making the capacitor susceptible to parasitic resonances in the 50 Mc. to
150 Mc. range of frequencies.
The question of optimum C/L ratiq and capacitor plate spacing is covered in Chapter
Thirteen. For all -band operation of a high power
stage, it is recommended that a capacitor just
large enough for 40 -meter phone operation be
chosen. (This will have sufficient capacitance
for phone operation on all higher frequency
bands.) Then use fixed padding capacitors for
operation on 80 meters. Such padding capacitors are available in air, ceramic, and vacuum
types.
Specially designed variable capacitors are
recommended for u -h -f work; ordinary capacitors often have "loops" in the metal frame
which may resonate near the operating frequency.
Variable vacuum capacitors
because of their small physical size have less inherent inductance per unit of capacity than do
variable air capacitors. Their losses are extremely low, and their dielectric strength is
high. Because of increased production the
cost of such units is now within the reach of
the designer of amateur equipment, and their
use is highly recommended in high power tank
Variable Vacuum
Capacitors
circuits.
18 -3
THE
Transmitter Design
RADIO
Tinned or stranded wire will show greater
losses at these frequencies. Tank coil and
tank capacitor leads should be of heavier wire
than other r -f leads.
The best type of flexible lead from the envelope of a tube to a terminal is thin copper
strip, cut from thin sheet copper. Heavy, rigid
leads to these terminals may crack the envelope glass when a tube heats or cools.
Wires carrying only a.f. or d.c. should be
chosen with the voltage and current in mind.
Some of the low- filament- voltage transmitting
tubes draw heavy current, and heavy wire must
be used to avoid voltage drop. The voltage is
low, and hence not much insulation is required.
Filament and heater leads are usually twisted
together. An initial check should be made on
the filament voltage of all tubes of 25 watts
or more plate dissipation rating. This voltage
should be measured right at the tube sockets.
If it is low, the filament transformer voltage
should be raised. If this is impossible, heavier
or parallel wires should be used for filament
leads, cutting down their length if possible.
Coaxial cable may be used for high voltage
leads when it is desirable to shield them from
r -f fields. RG -8 /U cable may be used at d -c
potentials up to 8000 volts, and the lighter
RG -17 /U may be used to potentials of 3000
volts. Spark -plug type high- tension wire may
be used for unshielded leads, and will withstand 10,000 volts.
If this cable is used, the high- voltage leads
Wire and Inductors
Any length of wire, no matter how short,
has a certain value of inductance. This property is of great help in making coils and inductors, but may be of great hindrance when
it is not taken into account in circuit design
and construction. Connecting circuit elements
(themselves having residual inductance) together with a conductor possessing additional
inductance can often lead to puzzling difficulties. A piece of no. 10 copper wire ten inches
long (a not uncommon length for a plate lead
in a transmitter) can have a self- inductance of
0.15 microhenries. This inductance and that
of the plate tuning capacitor together with the
plate -to- ground capacity of the vacuum tube
can form a resonant circuit which may lead to
parasitic oscillations in the v -h -f regions. To
keep the self- inductance at a minimum, all r -f
carrying leads should be as short as possible
and should be made out of as heavy material
as possible.
At the higher frequencies, solid enamelled
copper wire is most efficient for r -f leads.
may be cabled with filament and other low voltage leads. For high -voltage leads in low poc/ r exciters, where the plate voltage is not
over 450 volts, ordinary radio hookup wire of
good quality will serve the purpose.
No r -f leads should be cabled; in fact it is
better to use enamelled or bare copper wire
for r -f leads and rely upon spacing for insulation. All r -f joints should be soldered, and the
joint should be a good mechanical junction
before solder is applied.
The efficiency and Q of air coils commonly
used in amateur equipment is a factor of the
shape of the coil, the proximity of the coil to
other objects (including the coil form) and the
material of which the coil is made. Dielectric
losses in so- called "air wound" coils are
low and the Q of such coils runs in the neighborhood of 300 to 500 at medium frequencies.
Unfortunately, most of the transmitting type
plug -in coils on the market designed for link
coupling have far too small a pick up link for
proper operation at 7 Mc. and 3.5 Mc. The coefficient of coupling of these coils is about
0.5, and additional means must be employed
to provide satisfactory coupling at these low
frequencies. Additional inductance in series
with the pick up link, the whole being reso-
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HANDBOOK
Rc
Inductors
L
Rc
357
L
"°-z'
Rc
L
C
¡_
C DISTRIBUTED
Figure
ELECTRICAL
EQUIVALENT OF
R -F
nated to the operating frequency will often
permit satisfactory coupling.
Coil Placement
For best Q a coil should be
in the form of a solenoid with
length from one to two times the diameter. For
minimum interstage coupling, coils should be
made as small physically as is practicable.
The coils should then be placed so that adjoining coils are oriented for minimum mutual
coupling. To determine if this condition exists,
apply the following test: the axis of one of the
two coils must lie in the plane formed by the
center turn of the other coil. If this condition
is not met, there will be appreciable coupling
unless the unshielded coils are very small in
diameter or are spaced a considerable distance from each other.
Insulation
On frequencies above 7 Mc., cera-
mic, polystyrene, or blycalex insulation is to be recommended. Cold flow must
be considered when using polystyrene (Amphenol 912, etc.). Bakelite has low losses on
the lower frequencies but should never be
used in the field of high- frequency tank circuits.
Lucite (or Plexiglas), which is available
in rods, sheets, or tubing, is satisfactory for
use at all radio frequencies where the r-J voltages are not especially high. It is very easy
to work with ordinary tools and is not expensive. The loss factor depends to a considerable extent upon the amount and kind of plas-
ticizer used.
The most important thing to keep in mind
regarding insulation is that the best insulation
is air. If it is necessary to reinforce air -wound
coils to keep turns from vibrating or touching,
use strips of Lucite or polystyrene cemented
in place with Amphenol 912 coil dope. This
will result in lower losses than the commonly
used celluloid ribs and Duco cement.
8
CHOKE AT VARIOUS FREQUENCIES
At low frequencies, the distributed capacity
has little effect and the electrical equivalent
circuit of the r -f choke is as shown in figure
8A. As the operating frequency of the choke
is raised the effect of the distributed capacity
becomes more evident until at some particular
frequency the distributed capacity resonates
with the inductance of the choke and a parallel
resonant circuit is formed. This point is shown
in figure 8B. As the frequency of operation
is further increased the overall reactance of
the choke becomes capacitive, and finally a
point of series resonance is reached (figure
8C.). This cycle repeats itself as the operating
frequency is raised above the series resonant
point, the impedance of the choke rapidly becoming lower on each successive cycle. A
chart of this action is shown in figure 9. It
can be seen that as the r -f choke approaches
and leaves a condition of series resonance,
the performance of the choke is seriously impaired. The condition of series resonance
may easily be found by shorting the terminals
of the r -f choke in question with a piece of
wire and exploring the windings of the choke
with a grid-dip oscillator. Most commercial
transmitting type chokes have series resonances in the vicinity of 11 Mc. or 24 Mc.
p.o :ill=
,
/
5
R -f
chokes may be considered to be special inductances designed to have a
high value of impedance over a large range of
frequencies. A practical r -f choke has inductance, distributed capacitance, and resistance.
Radio Frequency
Chokes
E
111111111V".
1111111/11116/11
/Iv
IS
20
25
FREQUENCY (MC.)
Figure 9
FREQUENCY- IMPEDANCE CHARACTERISTICS FOR TYPICAL PIE -WOUND
R -F CHOKES
www.americanradiohistory.com
358
LEAD
NDUCTANCE
Figure
10
GROUND LOOPS IN AMPLIFIER STAGES
Using chassis return
8. Common ground point
A.
18 -4
Grounds
At frequencies of 30 Mc. and below, a chasmay be considered as a fixed ground refer-
sis
THE
Transmitter Design
ence, since its dimensions are only a fraction
of a wavelength. As the frequency is increased
above 30 Mc., the chassis must be considered
as a conducting sheet on which there are
points of maximum current and potential. However, for the lower amateur frequencies, an
object may be assumed to be at ground potential when it is affixed to the chassis.
In transmitter stages, two important current
loops exist. One loop consists of the grid circuit and chassis return, and the other loop
consists of the plate circuit and chassis return. These two loops are shown in figure 10A.
It can be seen that the chassis forms a return
for both the grid and plate circuits, and that
ground currents flow in the chassis towards
the cathode circuit of the stage. For some
years the theory has been to separate these
ground currents from the chassis by returning
all ground leads to one point, usually the
cathode of the tube for the stage in question.
This is well and good if the ground leads are
of minute length and do not introduce cross
couplings between the leads. Such a technique
is illustrated in figure 1013. wherein all stage
components are grounded to the cathode pin
of the stage socket. However, in transmitter
RADIO
construction the physical size of the components prevent such close grouping. It is necessary to spread the components of such a
stage over a fairly large area. In this case it
is best to ground items directly to the chassis
at the nearest possible point, with short, direct
grounding leads. The ground currents will
flow from these points through the low inductance chassis to the cathode return of the
stage. Components grounded on the top of the
chassis have their ground currents flow through
holes to the cathode circuit which is usually
located on the bottom of the chassis, since
such currents travel on the surface of the chassis. The usual "top to bottom" ground path
is through the hole cut in the chassis for the
tube socket. When the gain per stage is relatively low, or there are only a small number
of stages on a chassis this universal grounding system is ideal. It is only in high gain
stages (i -f strips) where the "gain per inch"
is very high that circulating ground currents
will cause operational instability.
Intercoupling of It is important to prevent inGround Currents tercoupling of various different ground currents when the
chassis is used as a common ground return.
To keep this intercoupling at a minimum, the
stage should be completely shielded. This
will prevent external fields from generating
spurious ground currents, and prevent the
ground currents of the stage from upsetting
the action of nearby stages. Since the ground
currents travel on the surface of the metal,
the stage should be enclosed in an electrically
tight box. When this is done, all ground currents generated inside the box will remain in
the box. The only possible means of escape
for fundamental and harmonic currents are imperfections in this electrically tight box. Whenever we bring a wire lead into the box, make
a ventilation hole, or bring a control shaft
through the box we create an imperfection. It
is important that the effect of these imperfections be reduced to a minimum.
18 -5
Holes, Leads and Shafts
Large size holes for ventilation may be put
in an electrically tight box provided they are
properly screened. Perforated metal stock having many small, closely spaced holes is the
best screening material. Copper wire screen
may be used provided the screen wires are
bonded together every few inches. As the wire
corrodes, an insulating film prevents contact
between the individual wires, and the attenuation of the screening suffers. The screening
material should be carefully soldered to the
www.americanradiohistory.com
HANDBOOK
359
Shielding
EXTERNAL FIELD
TIN CAN BOTTOM WITH
FLUTED EDGE PRESSED
AGAINST PANEL
HOLES FOR
METER STUDS
RUBBER GROMMET
METER NUT
COAXIAL SOCKET
COAXIAL PLUG
.001 CERAMIC
RFC
,OOI
PANEL
PANEL
METER
METER
LEAD
CERAMIC
PLUTEO EDGES TO MAKE
000D ELECTRICAL CONTACT WITH PANEL
CENTER
CONDUCTOR
.
RIGHT
Figure 11A
SIMPLE METER SHIELD
-OREN- BOX
HOLE
box, or bolted with a spacing of not less than
two inches between bolts. Mating surfaces of
the box and the screening should be clean.
A screened ventilation opening should be
roughly three times the size of an equivalent
unscreened opening, since the screening represents about a 70 per cent coverage of the
area. Careful attention must be paid to equipment heating when an electrically tight box
is used.
Commercially available panels having half inch ventilating holes may be used as part of
the box. These holes have much less attenuation than does screening, but will perform in a
satisfactory manner in all but the areas of
weakest TV reception. If it is desired to reduce leakage from these panels to a minimum,
the back of the grille must be covered with
screening tightly bonded to the panel.
Doors may be placed in electrically tight
boxes provided there is no r -f leakage around
the seams of the door. Electronic weatherstripping or metal "finger stock" may be used
to seal these doors. A long, narrow slot in a
closed box has the tendency to act as a slot
antenna and harmonic energy may pass more
readily through such an opening than it would
through a much larger circular hole.
Variable capacitor shafts or switch shafts
may act as antennas, picking up currents inside the box and re- radiating them outside of
the box. It is necessary either to ground the
shaft securely as it leaves the box, or else to
make the shaft of some insulating material.
A two or three inch panel meter requires a
large leakage hole if it is mounted in the wall
of an electrically tight box. To minimize leak age, the meter leads should be by- passed and
shielded. The meter should be encased with a
metal shield that makes contact to the box
entirely around the meter. The connecting
studs of the meter may project through the
back of the metal shield. Such a shield may
be made out of the end of a tin can of correct
EXTERNAL
FIELD
INTERNAL GROUND
J
CURRENTS
[[
LL
((
OÑ EXTE0.1EORÓF BO1AlL
CURRENTS
WRONG
Figure 11B
of coaxial connectors on electrically
tight box prevents escape of ground currents
from interior of box. At the same time external fields are not conducted into the interior
of the box.
Use
diameter, cut to fit the depth of the meter.
This complete shield assembly is shown in
figure 11A.
Careful attention should be paid to leads
entering and leaving the electrically tight
box. Harmonic currents generated within the
box can easily flow out of the box on power
or control leads, or even on the outer shields
of coaxially shielded wires. Figure 11B illustrates the correct method of bringing shielded
cables into a box where it is desired to preserve the continuity of the shielding.
Unshielded leads entering the box must be
carefully filtered to prevent fundamental and
harmonic energy from escaping down the lead.
Combinations of r -f chokes and low inductance
by -pass capacitors should be used in power
leads. If the current in the lead is high, the
chokes must be wound of large gauge wire.
Composition resistors may be substituted for
the r -f chokes in high impedance circuits.
Bulkhead or feed - through type capacitors are
preferable when passing a lead through a
shield partition. A summary of lead leakage
with various filter arrangements is shown in
figure 12.
Internal Leads Leads that connect two points
within an electrically tight box
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TEST
FIELD STRENGTH
IN UV
NO.
12000
I
THE
Transmitter Design
360
2
10000
3
630
B
SHIELDED OSCILLATOR
,o-SMALL
HOLE IN SHIELD
TO
OSC.
I
CI
WELDED HOOK-UP WIRE
C2 RI
600
_2
150
S
6
70
7
140
R
I
z
600
110
10
c.
C
RFC
RiC
C4 RFC
25
TRACE
12
RFC
I
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x
L--_1---- r-i
J.c3
- MOOR CARBON
RFC-OHMITE 2-50
R
CI
SO
75MLFCERAMIC
FEED-THROUGH
4 RFC
IEID
_
D
WZRE
C2 -.005 DISC CERAMIC
C3 - .01 SPRAGUE HI -PASS
C4 - 005 CERAMIC
FEED -THROUGH
Figure 12
LEAD LEAKAGE WITH VARIOUS
LEAD FILTERING SYSTEMS
(COURTESY WIDBM)
may pick up fundamental and harmonic currents if they are located in a strong field of
flux. Any lead forming a closed loop with itself will pick up such currents, as shown in
figure 13. This effect is enhanced if the lead
happens to be self -resonant at the frequency
at which the exciting energy is supplied. The
solution for all of this is to by -pass all internal power leads and control leads at each
end, and to shield these leads their entire
length. All filament, bias, and meter leads
should be so treated. This will make the job
of filtering the leads as they leave the box
much easier, since normally "cool" leads
within the box will not have picked up spurious currents from nearby "hot" leads.
was operated near this frequency marked instability was noted, and the filaments of the
810 tubes increased in brilliance when plate
voltage was applied to the amplifier, indicating the presence of r.f. in the filament circuit.
Changing the filament by -pass capacitors to
.01-pfd. lowered the filament resonance frequency to 2.2 Mc. and cured this effect. A
ceramic capacitor of .01 -pfd. used as a filament by -pass capacitor on each filament leg
seems to be satisfactory from both a resonant
and a TVI point of view. Filament by -pass
capacitors smaller in value than .01-pfd.
should be used with caution.
Various parasitic resonances are also found
in plate and grid tank circuits. Push -pull tank
circuits are prone to double resonances, as
shown in figure 14. The parasitic resonance
circuit is usually several megacycles higher
than the actual resonant frequency of the full
tank circuit. The cure for such a double resonance is the inclusion of an r -f choke in the
center tap lead to the split coil.
From a point of view of electrical properties, aluminum
is a poor chassis material. It is difficult to
make a soldered joint to it, and all grounds
must rely upon a pressure joint. These presChassis Material
Figure
13
SHIELDED
SHIELDED
COMPARTMENT
COMPARTMENT
RADIATION
FIELD
\
\.
HOLE
ICKU'
LOOP
RE-RADIATED
FIELD
1
BY-PASS
CAPACITOR
BY -PASS
CAPACITOR
ILLUSTRATION OF
HOW A SUPPOSEDLY
GROUNDED POWER LEAD CAN COUPLE
ENERGY FROM ONE COMPARTMENT
TO ANOTHER
OMPARTMENT
Parasitic Resonances
Filament leads within vacuum tubes may
resonate with the filament by -pass capacitors
at some particular frequency and cause instability in an amplifier stage. Large tubes of
the 810 and 250TH type are prone to this spurious effect. In particular, a push -pull 810 amplifier using .001 -µfd. filament by -pass capacitors had a filament resonant loop that fell in
the 7 -Mc. amateur band. When the amplifier
WRONG
RADIATION
LOOP
LECTRICALLY -TIGHT
18 -6
RADIO
\\
RADIATION
FIELD
R
IGHT
ELECTRICALLY -TIGHT
COMPARTMENT
BULKHEAD TYPE
fBY -PASS CAPACITOR
ILLUSTRATION OF LEAD ISOLATION BY
PROPER USE OF BULKHEAD BYPASS
CAPACITOR
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Parasitic Oscillations
HANDBOOK
RIGHT
WRONG
Figure
361
final amplifier stage that might be very severe
if the plate voltage were left on and the excitation were keyed.
In some cases, an all -wave receiver will
prove helpful in locating v -h -f spurious oscillations, but it may be necessary to check from
several hundred megacycles downward in frequency to the operating range. A normal harmonic is weaker than the fundamental but of
good tone; a strong harmonic or a rough note
at any frequency generally indicates a para-
sitic.
14
DOUBLE RESONANCE EFFECTS IN PUSH PULL TANK CIRCUIT MAY BE ELIMINATED BY THE INSERTION OF ANY
R -F CHOKE IN THE COIL CENTER
TAP LEAD
In general, the cure for parasitic oscillation
is two-fold: The oscillatory circuit is damped
until sustained oscillation is impossible, or
it is detuned until oscillation ceases. An examination of the various types of parasitic oscillations and of the parasitic oscillatory circuits will prove handy in applying the correct
cure.
later
date because of high resistivity caused by
the formation of oxides from eletrolytic action
in the joint. However, the ease of working
sure joints are prone to give trouble at a
and forming the aluminum material far outweighs the electrical shortcomings, and aluminum chassis and shielding may be used
with good results provided care is taken in
making all grounding connections. Cadmium
and zinc plated chassis are preferable from a
corrosion standpoint, but are much more difficult to handle in the home workshop.
18 -7
Parasitic Oscillation
in R -F Amplifiers
Parasitics (as distinguished from sell- oscillation on the normal tuned frequency of the
amplifier) are undesirable oscillations either
of very high or very low frequencies which
may occur in radio -frequency amplifiers.
They may cause spurious signals (which
are often rough in tone) other than normal harmonics, hash on each side of a modulated carrier, key clicks, voltage breakdown or flashover, instability or inefficiency, and shortened
life or failure of the tubes. They may be damped
and stop by themselves after keying or modulation peaks, or they may be undamped and
build up during ordinary unmodulated transmission, continuing if the excitation is removed. They may result from series or parallel resonant circuits of all types. Due to neutralizing lead length and the nature of most
parasitic circuits, the amplifier usually is not
neutralized for the parasitic frequency.
Sometimes the fact that the plate supply is
keyed will obscure parasitic oscillations in a
Low Frequency
One type of unwanted
oscillation often occurs
in shunt -fed circuits in
which the grid and plate chokes resonate, coupled through the tube's interelectrode capacitance. This also can happen with series feed.
This oscillation is generally at a much lower
frequency than the operating frequency and
will cause additional carriers to appear, spaced
Parasitic Oscillations
from perhaps twenty to a few hundred kilocycles on either side of the main wave. Such a
circuit is illustrated in figure 15. In this case,
RFC, and RFC2 form the grid and plate inductances of the parasitic oscillator. The neutralizing capacitor, no longer providing out -ofphase feedback to the grid circuit actually enhances the low frequency oscillation. Because
of the low Q of the r -f chokes, they will usually run warm when this type of parasitic oscillation is present and may actually char and
burn up. A neon bulb held near the oscillatory
circuit will glow a bright yellow, the color
appearing near the glass of the neon bulb and
not between the electrodes.
One cure for this type of oscillation is to
change the type of choke in either the plate
or the grid circuit. This is a marginal cure,
because the amplifier may again break into the
same type of oscillation when the plate voltage is raised slightly. The best cure is to remove the grid r -f choke entirely and replace
it with a wirewound resistor of sufficient watt-
age to carry the amplifier grid current. If the
inclusion of such a resistor upsets the operating bias of the stage, an r -f choke may be
used, with a 100 -ohm 2 -watt carbon resistor
in series with the choke to lower the operating
Q of the choke. If this expedient does not
eliminate the condition, and the stage under
investigation uses a beam -tetrode tube, negative resistance can exist in the screen circuit
www.americanradiohistory.com
THE
Transmitter Design
362
RADIO
2
RFC2
RFC
R
F.
RFC!
RFCz
GRID
vLATE
TANIÇ
CURE
PARASITIC CIRCUIT FOR
LOW FREQ. OSCILLATION
CIRCUIT
Figure
THE CAUSE
AND CURE
15
OF LOW
of such tubes. Try larger and smaller screen
by -pass capacitors to determine whether or not
they have any effect. If the condition is corning from the screen circuit an audio choke with
a resistor across it in series with the screen
feed lead will often eliminate the trouble.
Low -frequency parasitic oscillations can
often take place in the audio system of an AM
transmitter, and their presence will not be
known until the transmitter is checked on a
receiver. It is easy to determine whether or
not the oscillations are coming from the modulator simply by switching off the modulator
tubes. If the oscillations are coming from the
modulator, the stage in which they are being
generated can be determined by removing tubes
successively, starting with the first speech
amplification stage, until the oscillation stops.
When the stage has been found, remedial steps
can be taken on that stage.
If the stage causing the oscillation is a lowlevel speech stage it is possible that the
trouble is coming from r -f or power- supply
feedback, or it may be coming about as a result of inductive coupling between two transformers. If the oscillation is taking place in
a high -level audio stage, it is possible that
inductive or capacitive coupling is taking
place back to one of the low -level speech
stages. It is also possible, in certain cases,
that parasitic push -pull oscillation can take
place in a Class B or Class AB modulator as
a result of the grid -to-plate capacitance within the tubes and in the stage wiring. This condition is more likely to occur if capacitors
have been placed across the secondary of the
driver transformer and across the primary of
the modulation transformer to act in the reduction of the amplitude of the higher audio frequencies. Relocation of wiring or actual neutralization of the audio stage in the manner
used for r -f stages may be required.
FREQUENCY PARASITICS
It may be said in general that the presence
low- frequency parasitics indicates that
somewhere in the oscillating circuit there is
an impedance which is high at a frequency in
the upper audio or low r -f range. This impedance may include one or more r -f chokes of
the conventional variety, power supply chokes,
modulation components, or the high impedance
may be presented simply by an RC circuit
such as might be found in the screen -feed circuit of a beam -tetrode amplifier stage.
of
18 -8
Elimination of V -H -F
Parasitic Oscillations
V -h -f parasitic oscillations are often difficult to locate and difficult to eliminate since
their frequency often is only moderately above
the desired frequency of operation. But it may
be said that v -h -f parasitics always may be
eliminated if the operating frequency is appreciably below the upper frequency limit for the
tubes used in the stage. However, the elimination of a persistent parasitic oscillation on
a frequency only moderately higher than the
desired operating frequency will involve a
sacrifice in either the power output or the
power sensitivity of the stage, or in both.
Beam- tetrode stages, particularly those
using 807 type tubes, will almost invariably
have one or more v -h -f parasitic oscillations
unless adequate precautions have been taken
in advance. Many of the units described in
the constructional section of this edition had
parasitic oscillations when first constructed.
But these oscillations were eliminated in each
case; hence, the expedients used in these
equipments should be studied. V-h -f parasitics
may be readily identified, as they cause a
www.americanradiohistory.com
Parasitic Oscillations
HANDBOOK
363
neon lamp to have a purple glow close to the
electrodes when it is excited by the parasitic
energy.
Parasitic Oscillations
with Triodes
Triode stages are less
subject to parasitic os-
cillations primarily because of the much lower power sensitivity of
such tubes as compared to beam tetrodes. But
such oscillations can and do take place. Usually, however, it is not necessary to incorporate
losser resistors as normally is the case with
beam tetrodes, unless the triodes are operated
quite near to their upper frequency limit, or
the tubes are characterized by a relatively
high transconductance. Triode v -h -f parasitic
oscillations normally may be eliminated by adjustment of the lengths and effective inductance of the leads to the elements of the tubes.
In the case of triodes, v -h -f parasitic oscillations often come about as a result of inductance in the neutralizing leads. This is particularly true in the case of push -pull amplifiers.
The cure for this effect will usually be found
in reducing the length of the neutralizing leads
and increasing their diameter. Both the reduction in length and increase in diameter will
reduce the inductance of the leads and tend
to raise the parasitic oscillation frequency
until it is out of the range at which the tubes
will oscillate. The use of straightforward cir-
cuit design with short leads will assist in
forestalling this trouble at the outset. Butterfly -type tank capacitors with the neutralizing
capacitors built into the unit (such as the
B &W type) are effective in this regard.
V-h -f parasitic oscillations may take place
as a result of inadequate by-passing or long
by -pass leads in the filament, grid- return and
plate -return circuits. Such oscillations also
can take place when long leads exist between
the grids and the grid tuning capacitor or between the plates and the plate tuning capacitor. The grid and plate leads should be kept
short, but the leads from the tuning capacitors
to the tank coils can be of any reasonable
length insofar as parasitic oscillations are
concerned. In an amplifier where oscillations
have been traced to the grid or plate leads,
their elimination can often be effected by making the grid leads much longer than the plate
leads or vice versa. Sometimes parasitic oscillations can be eliminated by using iron or
nichrome wire for the grid or plate leads, or
for the neutralizing leads. But in any event
it will always be found best to make the neutralizing leads as short and of as heavy conductor as is practicable.
In cases where it has been found that increased length in the grid leads for an amplifier is required, this increased length can often
be wound into the form of a small coil and still
Figure
16
GRID PARASITIC SUPPRESSORS IN PUSH PULL TRIODE STAGE
obtain the desired effect. Winding these small
coils of iron or nichrome wire may sometimes
be of assistance.
To increase losses at the parasitic frequency,
the parasitic coils may be wound on 100 -ohm
resistors. These "lossy" suppressors
should be placed in the grid leads of the tubes
close to the grid connection, as shown in fig2 -watt
ure 16.
Parasitics with
Where beam -tetrode tubes are
used in the stage which has
been found to be generating
the parasitic oscillation, all the foregoing
suggestions apply in general. However, there
Beam Tetrodes
are certain additional considerations involved
in elimination of parasitics from beam -tetrode
amplifier stages. These considerations involve
the facts that a beam -tetrode amplifier stage
has greater power sensitivity than an equivalent triode amplifier, such a stage has a certain amount of screen -lead inductance which
may give rise to trouble, and such stages have
a small amount of feedback capacitance.
Beam - tetrode stages often will require the
inclusion of a neutralizing circuit to eliminate
oscillation on the operating frequency. However, oscillation on the operating frequency
normally is not called a parasitic oscillation,
and different measures are required to eliminate the condition.
Basically, parasitic oscillations in beam tetrode amplifier stages fall into two classes:
cathode -grid- screen oscillations, and cathode screen -plate oscillations. Both these types of
oscillation can be eliminated through the use
of a parasitic suppressor in the lead between
the screen terminal of the tube and the screen
by -pass suppressor, as shown in figure 17.
Such a suppressor has negligible effect on the
by- passing effect of the screen at the operating frequency. The method of connecting this
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364
THE
Transmitter Design
PC=ST *18E.
RADIO
ON
52/1, 2W CAR-
BON RESISTOR
RFC=ONM/TE Z50 OR
EQUIVALENT
Figure 17
PARASITIC SUPPRESSION CIRCUIT FOR TETRODE TUBES
SCREEN
suppressor to tubes having dual screen leads
is shown in figure 18. At the higher frequencies at which parasitics occur, the screen is
no longer at ground potential. It is therefore
necessary to include an r -f choke by-pass condenser filter in the screen lead after the parasitic suppressor. The screen lead, in addition,
should be shielded for best results.
During parasitic oscillations, considerable
r -f voltage appears on the screen of a tetrode
tube, and the screen by -pass condenser can
easily be damaged. It is best, therefore, to
employ screen by -pass condensers whose d -c
working voltage is equal to twice the maximum
applied screen voltage.
The grid- screen oscillations may occasionally be eliminated through the use of a parasitic suppressor in series with the grid lead
of the tube. The screen plate oscillations may
also be eliminated by inclusion of a parasitic
suppressor in series with the plate lead of the
tube. A suitable grid suppressor may be made
of a 22 -ohm 2 -watt Ohmite or Allen- Bradley
resistor wound with 8 turns of no. 18 enameled
wire. A plate circuit suppressor is more of a
problem, since it must dissipate a quantity of
power that is dependent upon just how close
the parasitic frequency is to the operating frequency of the tube. If the two frequencies are
close, the suppressor will absorb some of the
fundamental plate circuit power. For kilowatt
stages operating no higher than 30 Mc. a satisfactory plate circuit suppressor may be made
of five 570 -ohm 2 -watt carbon resistors in parallel, shunted by 5 turns of no. 16 enameled
wire, % inch diameter and % inch long (figure
19A and B).
The parasitic suppressor for the plate circuit of a small tube such as the 5763, 2E26,
807, 6146 or similar type normally may consist of a 47 -ohm carbon resistor of 2 -watt size
with 6 turns of no. 18 enameled wire wound
around the resistor. However, for operation
above 30 Mc., special tailoring of the value
lj
""``""ftmeYrnrw.
Figure 18
APPLICATION OF SCREEN
PARASITIC SUPPRESSION CIRCUIT
PHOTO
OF
OF FIGURE 17
of the resistor and the size of the coil wound
around it will be required in order to attain
satisfactory parasitic suppression without excessive power loss in the parasitic suppressor.
Isolation between the grid
and plate circuits of a tetrode tube is not perfect. For maximum stability, it is recommended that the tetrode stage
be neutralized. Neutralization is absolutely
necessary unless the grid and plate circuits
Tetrode Screening
of the tetrode stage are each completely isolated from each other in electrically tight
boxes. Even when this is done, the stage will
show signs of regeneration when the plate
and grid tank circuits are tuned to the same
frequency. Neutralization will eliminate this
regeneration. Any of the neutralization circuits described in the chapter Generation of
R -F Energy may be used.
18 -9
Checking for -Parasitic
Oscillations
It is an unusual transmitter which harbors
no parasitic oscillations when first constructed
www.americanradiohistory.com
Parasitic Oscillations
HANDBOOK
365
PC
PC
FOR
PC
=! T
PC
R/eE. ON
22d.2W. COM-
507,
ETC.
=ArI /IE.ON47/1,2W
POSITION RESISTOR
COMPOSITION RESISTOR
FOR 4
PC =
S
-250A,
ETC.
-570!2, 2W COMPOSITION
RESISTORS IN PARALLEL WITH
ST.
0 /eE.
//4.
O
Figure
19
PLATE AND GRID PARASITIC SUPPRESSION IN TETRODE TUBES
and tested. Therefore it is always wise to follow a definite procedure in checking a new
transmitter for parasitic oscillations.
Parasitic oscillations of all types are most
easily found when the stage in question is
running by itself, with full plate (and screen)
voltage, sufficient protective bias to limit the
plate current to a safe value, and no excitation. One stage should be tested at a time,
and the complete transmitter should never be
put on the air until all stages have been thoroughly checked for parasitics.
To protect tetrode tubes during tests for
parasitics, the screen voltage should be applied through a series resistor which will limit
the screen current to a safe value in case the
plate voltage of the tetrode is suddenly removed when the screen supply is on. The correct procedure for parasitic testing is as follows (figure 20):
1. The stage in question should be coupled
to a dumpy load, and tuned up in correct oper-
ating shape. Sufficient protective bias should
be applied to the tube at all times. For protection of the stage under test, a lamp bulb
should be added in series with one leg of the
primary circuit of the high voltage power supply. As the plate supply load increases during
a period of parasitic oscillation, the voltage
drop across the lamp increases, and the effective plate voltage drops. Bulbs of various
size may be tried to adjust the voltage under
testing conditions to the correct amount. If a
Variac or Powerstat is at hand, it may be used
in place of the bulbs for smoother voltage control. Don't test for parasitics unless some type
of voltage control is used on the high voltage
supply! When a stage breaks into parasitic
oscillations, the plate current increases violently, and some protection to the tube under
test must be used.
2. The r -f excitation to the tube should now
be removed. When this is done, the grid, screen
and plate currents of the tube should drop to
zero. Grid and plate tuning condensers should
be tuned to minimum capacity. No change in
resting grid, screen or plate current should
be observed. If a parasitic is present, grid current will flow, and there will be an abrupt increase in plate current. The size of the lamp
bulb in series with the high voltage supply may
be varied until the stage can oscillate contin-
uously, without exceeding the rated plate dissipation of the tube.
3. The frequency of the parasitic may now
be determined by means of an absorption wave
meter, or a neon bulb. Low frequency oscillations will cause a neon bulb to glow yellow.
High frequency oscillations will cause the
bulb to have a soft, violet glow. Once the frequency of oscillation is determined, the cures
suggested in this chapter may be applied to
the stage.
4. When the stage can pass the above test
with no signs of parasitics, the bias supply of
the tube in question should be decreased until
the tube is dissipating its full plate rating
when full plate voltage is applied, with no r-f
TO BE TESTED
FOR
EXCITER CONTROL
SWITCH
nl
AMPLIFIER STAGE
EXCITER
PARASITICS
DUMMY
LOAD
°
BIAS SUPPLY
HIGH VOLTAGE
POWER SUPPLY
VARIAC OR
LIGHT BULBS
Figure 20
SUGGESTED TEST SETUP FOR PARASITIC
TESTS
www.americanradiohistory.com
4
A.C.
SUPPLY
366
Transmitter
Design
excitation. Excitation may now be applied and
the stage loaded to full input into a dummy
load. The signal should now be monitored in
a nearby receiver which has the antenna terminals grounded or otherwise shorted out. A
series of rapid dots should be sent, and the
frequency spectrum for several megacycles
each side of the carrier frequency carefully
searched. If any vestige of parasitic is left,
it will show up as an occasional "pop" on a
keyed dot. This "pop" may be enhanced by a
slight detuning of either the grid or plate circuit.
5. If such a parasitic shows up, it means
that the stage is still not stable, and further
measures must be applied to the circuit. Parasitic suppressors may be needed in both screen
and grid leads of a tetrode, or perhaps in both
grid and neutralizing leads of a triode stage.
As a last resort, a 10,000 -ohm 25 -watt wire wound resistor may be shunted across the grid
coil, or grid tuning condenser of a high powered stage. This strategy removed a keying
pop that showed up in a commercial transmitter, operating at a plate voltage of 5000.
Test for Parasitic
Tendency in Tetrodo
Amplifiers
is common experience
to develop an engineer ing model of a new
It
equipment that is apparently free of parasitics and then find troublesome oscillations showing up in production
units. The reason for this is that the equipment
has a parasitic tendency that remains below
the verge of oscillation until some change in
a component, tube gain, or operating condition
raises the gain of the parasitic circuit enough
to start oscillation.
In most high frequency transmitters there
are a great many resonances in the tank circuits at frequencies other than the desired
SIGNAL GENERATOR
100CC -20O MC
Figure
21
PARASITIC GAIN MEASUREMENT
Grid -dip oscillator and vacuum tube
voltmeter may be used to measure para-
sitic
stage
gain over
region.
IOOkc -200mc
operating frequency. Most of these parasitic
resonant circuits are not coupled to the tube
and have no significant tendency to oscillate.
A few, however, are coupled to the tube in
some form of oscillatory circuit. If the regeneration is great enough, oscillation at the parasitic frequency results. Those spurious circuits
existing just below oscillation must be found
and suppressed to a safe level.
One test method is to feed a signal from a
grid -dip oscillator into the grid of a stage and
measure the resulting signal level in the plate
circuit of the stage, as shown in figure 21. The
test is made with all operating voltages applied
to the tubes. Class C stages should have bias
reduced so a reasonable amount of static plate
current flows. The grid -dip oscillator is tuned
over the range of 100 kc to 200 mc. and the
relative level of the r -f voltmeter is watched
and the frequencies at which voltage peaks
occur are noted. Each significant peak in voltage gain in the stage must be investigated. Circuit changes or suppression must then be added
to reduce all peaks by 10 db or more in amplitude.
www.americanradiohistory.com
CHAPTER NINETEEN
Television and Broadcast
Interference
The problem of interference to television
reception is best approached by the philosophy discussed in Chapter Eighteen. By correct
design procedure, spurious harmonic generation in low frequency transmitters may be held
to a minimum. The remaining problem is twofold: to make sure that the residual harmonics
generated by the transmitter are not radiated,
and to make sure that the fundamental signal
of the transmitter does not overload the television receiver by reason of the proximity of
one to the other.
In an area of high TV- signal field intensity
the TVI problem is capable of complete solution with routine measures both at the amateur
transmitter and at the affected receivers. But
in fringe areas of low TV- signal field strength
the complete elimination of TVI is a difficult
and challenging problem. The fundamentals
illustrated in Chapter Fifteen must be closely
followed, and additional antenna filtering of
the transmitter is required.
19 -1
Even if the amateur transmitter
were perfect and had no harmonic radiation or spurious
emissions whatever, it still would be likely to
cause overloading of TV sets whose antennas
were within a few hundred feet of the transmitting antenna. This type of overloading is
essentially the same as the common type of
BCI encountered when operating a medium power or high -power amateur transmitter within a few hundred feet of the normal type of
BCL receiver. The field intensity in the immediate vicinity of the transmitting antenna
is sufficiently high that the amateur signal
will get into the BC or TV set either through
overloading of the front end, or through the
i -f, video, or audio system. A characteristic
TV Set
Overloading
of this type of interference is that it always
will be eliminated when the transmitter temporarily is operated into a dummy antenna.
Another characteristic of this type of overloading is that its effects will be substantially continuous over the entire frequency
coverage of the BC or TV receiver. Channels
2 through 13 will be affected in approximately
the same manner.
With the overloading type of interference
the problem is simply to keep the fundamental
of the transmitter out of the affected receiver.
Other types of interference may or may not
show up when the fundamental is taken out of
the TV set (they probably will appear), but at
least the fundamental must be eliminated first.
The elimination of the transmitter fundamental from the TV set is normally the only operation performed on or in the vicinity of the TV
receiver. After the fundamental has been elimi-
Types of Television
Interference
There are three main types of TVI which
may be caused singly or in combination by the
emissions from an amateur transmitter. These
types of interference are:
(1) Overloading of the TV set by the trans(2)
(3)
mitter fundamental
Impairment of the picture by spurious
emissions
Impairment of the picture by the radiation of harmonics .
367
www.americanradiohistory.com
368
TV and Broadcast Interference
THE
SHIELD BOX
CI
(SNORT
0
300-OHM
TO
LINE FROM
INPUT
S
ANTENNA
OF TV
I
SE
LEADS
300 OHM
LINE FROM
ANTENNA
_
L2
pA
FOR
COAx
FIiTING
Jl
ANTENNA
TERMINALS
OF TV SET
TO
Figure
TUNED TRAPS FOR THE TRANSMITTER
FUNDAMENTAL
The arrangement at (A) has proven to be effective in eliminating the condition of general blocking as caused by a 28 -Mc. transmitter in the vicinity of a TV receiver. The
tuned circuits LI -CI are resonated separately to the frequency of transmission. The adjustment may be done at the station, or it
may be accomplished at the TV receiver by
tuning for minimum interference on the TV
screen.
FOR 50
-75
Shown at (B) is an alternative arrangement
a
series -tuned circuit across the anten-
naced as a source of interference to reception,
work may then be begun on or in the vicinity
of the transmitter toward eliminating the other
two types of interference.
less standard BCItype practice is most commonly used in t a k i n g out
fundamental interference. Wavetraps and filters are installed, and the antenna system may
or may not be modified so as to offer less response to the signal from the amateur transmitter. In regard to a comparison between
wavetraps and filters, the same considerations
apply as have been effective in regard to BCI
for many years; wavetraps are quite effective
when properly installed and adjusted, but they
Taking Out
the Fundamental
More or
TERM ON
TV SET
I
COAX
FITTING
3
OHM COAXIAL LINE
Figure
1
na terminals of the TV set. The tuned circuit should be resonated to the operating
frequency of the transmitter. This arrangement gives less attenuation of the interfering signal than that at (A); the circuit has
proven effective with interference from transmitters on the 50 -Mc. band, and with low power 28 -Mc. transmitters.
ANTENNA
C2
II-E--d --F
L3
©
Li
`
TO
300 -OHM LINE, SHIELDED OR UNSHIELDED
C2
TO TV ANTENNA
with
RADIO
2
HIGH -PASS TRANSMISSION LINE FILTERS
The arrangement at (A) will stop the passing
of all signals below about 45 Mc. from the
antenna transmission line into the TV set.
Coils LI ore each 1.2 microhenrys (17 turns
no. 24 enam. closewound on 1/2-inch dia. polystyrene rod) with the center tap grounded.
It will be found best to scrape, twist, and
solder the center tap before winding the coil.
The number of turns each side of the tap may
then be varied until the tap is in the exact
center of the winding. Coil L2 is 0.6 microhenry (12 turns no. 24 enam. closewound on
1/2-inch dia. polystyrene rod). The capacitors
should be about 16.5 Wald., but either 15 or
20 pµtd. ceramic capacitors will give satis-
factory results. A similar filter for coaxial
antenna transmission line is shown at (B).
Both coils should be 0.12 microhenry (7 turns
no. 18 enam. spaced to % inch on 1/2-inch dia.
polystyrene rod). Capacitors C2 should be
75 µµtd. midget ceramics, while C3 should
be a 40 -µµtd. ceramic.
must be readjusted whenever the band of operation is changed, or even when moving from
one extreme end of a band to the other. Hence,
wavetraps are not recommended except when
operation will be confined to a relatively narrow portion of one amateur band. However, figure 1 shows two of the most common signal
trapping arrangements.
High -Pass Filters
High -pass
filters
in the
antenna lead of the TV set
have proven to be quite sat i s factory as a
means of eliminating TVI of the overloading
type. In many cases when the interfering transmitter is operated only on the bands below
7.3 Mc., the use of a high -pass filter in the
antenna lead has completely eliminated all
www.americanradiohistory.com
HANDBOOK
Harmonic
Radiation
369
TVI. In some cases the installation of a high pass filter in the antenna transmission line
and an a -c line filter of a standard variety has
proven to be completely effective in eliminating the interference from a transmitter operat-
ing in one of the lower
bands.
In
frequency
amateur
general, it is suggested that commercial-
ly manufactured high-pass filters be purchased.
Such units are available from a number of manufacturers at a relatively moderate cost. However, such units may be home constructed;
suggested designs are given in figures 2 and
3. Types for use both with coaxial and with
balanced transmission lines have been shown.
In most cases the filters may be constructed
in one of the small shield boxes which are
now on the market. Input and output terminals
may be standard connectors, or the inexpensive type of terminal strips usually used on
BC and TV sets may be employed. Coaxial
terminals should of course be employed when
a coaxial feed line is used to the antenna. In
any event the leads from the filter box to the
TV set should be very short, including both
the antenna lead and the ground lead to the
box itself. If the leads from the box to the set
have much length, they may pick up enough
signal to nullify the effects of the high -pass
filter.
Blocking from
50-Me. Signals
Operation on the 50 -Mc. amateur band in an area where
channel 2 is in use for TV
imposes a special problem in the matter of
blocking. The input circuits of most TV sets
are sufficiently broad so that an amateur signal on the 50 -Mc. band will ride through with
little attenuation. Also, the normal TV antenna
will have a quite large response to a signal
in the 50 -Mc. band since the lower limit of
channel 2 is 54 Mc.
High -pass filters of the normal type simply
are not capable of giving sufficient attenuation to a signal whose frequency is so close
to the necessary pass band of the filter. Hence,
a resonant circuit element, as illustrated in
figure :, must be used to trap out the amateur
field at the input of the TV set. The trap must
be tuned or the section of transmission line
cut, if a section of line is to be used for a
particular frequency in the 50 -Mc. band.
This frequency will have to be near the lower
frequency limit of the 50 -Mc. band to obtain
adequate rejection of the amateur signal while
still not materially affecting the response of
the receiver to channel 2.
Elimination of
Spurious Emissions
All spurious e m i s s i o n s
from amateur transmitters
(ignoring harmonic signals
eliminated to corn-
for the time being) must be
Figure
3
SERIES -DERIVED HIGH -PASS FILTER
This filter is designed for use in the
300 -ohm transmission line from the TV
antenna to the TV receiver. Nominal cutoff frequency is 36 Mc. and maximum rejection is at about 29 Mc.
Ct,C6- 15 -µµfd. zero- coefficient ceramic
C2,C3,C4,C5-20 -i fd. zero -coefficint cramic
L1,Lt -2.0 µh. About
24 turns no. 28 d.c.c.
wound to ?éi on '4" diameter polystyrene
rod. Turns should be adjusted until the
coil resonates to 29 Mc. with the asociated 15- ptµfd. capacitor.
L2-0.66 i h., 14 turns no. 28 d.c.c. wound
to Se" on t/ " die. polystyrene rod. Adjust
turns to resonate externally to 20 Mc.
with an auxiliary 100- µ{tfd. capacitor
whose value is accurately known.
ply with FCC r e g u l a t i o n s. But in the past
many amateur transmitters have emitted spurious signals as a result of key clicks, parasities, and overmodulation transients. In most
cases the operators of the transmitters were
not aware of these emissions since they were
radiated only for a short distance and hence
were not brought to his attention. But with
one or more TV sets in the neighborhood it
is probable that such spurious signals will
be brought quickly to his attention.
19 -2
Harmonic Radiation
After any condition of blocking at the TV
receiver has been eliminated, and when the
transmitter is completely free of transients
and parasitic oscillations, it is probable that
TVI will be eliminated in certain cases. Certainly general interference should be eliminated, particularly if the transmitter is a well
designed affair operated on one of the lower
frequency bands, and the station is in a high signal TV area. But when the transmitter is
to be operated on one of the higher frequency
bands, and particularly in a marginal TV area,
the job of TVI -proofing will just have begun.
The elimination of harmonic radiation from
the transmitter is a difficult and tedious job
which must be done in an orderly manner if
completely satisfactory results are to be obtained.
www.americanradiohistory.com
370
THE
TV and Broadcast Interference
TRANSMITTER
FUNDAMENTAL
7.0
7
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TV I.F
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POSSIBLE INTERFERENCE
1
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Figure
U-H-F CHANNELS
4
HARMONICS OF THE AMATEUR BANDS
Shown are the harmonic frequency ranges of the amateur bands between 7 and 54 Mc., with the
TV channels (and TV i -f systems) which are most likely to receive interference from these harmonics. Under certain conditions amateur signals in the 1.8 and 3.5 Mc. bands con cause interference as a result of direct pickup in the video systems of TV receivers which are not ade-
quately shielded.
cross -hatch or herringbone pattern on
the TV screen. This same general type
of picture also will occur in the case of
a narrow -band FM signal either with or
First it is well to become familiar with the
TV channels presently assigned, with the TV
intermediate frequencies commonly used, and
with the channels which will receive interference from harmonics of the various amateur bands. Figures 4 and 5 give this information.
Even a short inspection of figures 4 and 5
will make obvious the seriousness of the interference which can be caused by harmonics
of amateur signals in the higher frequency
bands. With any sort of reasonable precautions
in the design and shielding of the transmitter
it is not likely that harmonics higher than the
6th will be encountered. hence the main offenders in the way of harmonic interference
will be those bands above 14 -Mc.
Nature of
Harmonic Interference
Investigations into the
nature of the interference caused by amateur signals on the TV screen, assuming that
blocking has been eliminated as described
earlier in this chapter, have revealed the following facts:
An unmodulated carrier, such as a c -w
1.
signal with the key down or an AM signal without m o d u l at ion, will give a
without modulation.
A relatively strong AM signal will give
in addition to the herringbone a very
serious succession of light and dark
bands across the TV picture.
3.
A moderate strength c -w signal without
transients, in the absence of overloading of the TV set, will result merely in
the turning on and off of the herringbone
on the picture.
To discuss condition (1) above, the herringbone is a result of the beat note between the
TV video carrier and the amateur harmonic.
Hence the higher the beat note the less obvious will be the resulting cross -hatch. Further, it has been shown that a much stronger
signal is required to produce a discernible
herringbone when the interfering harmonic is
as far away as possible from the video carrier, without running into the sound carrier.
Thus, as a last resort, or to eliminate the last
vestige of interference after all corrective
measures have been taken, operate the transmitter on a frequency such that the interfer2.
www.americanradiohistory.com
HANDBOOK
Harmonic
VIDEO
SOUND
U
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TV
'CHANNEL'
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HIGH BAND
Figure 5
FREQUENCIES OF THE V -H -F TV CHANNELS
Showing the frequency ranges of TV channels 2 through 13, with
the picture carrier and sound
carrier frequencies also shown.
ing harmonic will fall as far as possible from
the picture carrier. The worst possible interference to the picture from a continuous carrier will be obtained when the interfering signal is very close in frequency to the video
carrier.
Isolating
Throughout the testing procedure it will be necessary to
have some sort of indicating
device as a means of determining harmonic field intensities. The best
indicator for field intensities some distance
from the transmitting antenna will probably be
the TV receiver of some neighbor with whom
friendly relations are still maintained. This
person will then be able to give a check, occasionally, on the relative nature of the interference. But it will probably be necessary to
go and check yourself periodically the results
obtained, since the neighbor probably will not
be able to give any sort of a quantitative analysis of the progress which has been made.
An additional device for checking relatively high field intensities in the vicinity of the
transmitter will be almost a necessity. A simple crystal diode wavemeter, shown in figure
6 will accomplish this function. Also, it will
be very helpful to have a receiver, with an S
meter, capable of covering at least the 50 to
100 Mc. range and preferably the range to 216
Mc. This device may consist merely of the
station receiver and a simple converter using
the two halves of a 6J6 as oscillator and
mixer.
the Source of
the Interference
The first check can best be made with the
neighbor who is receiving the most serious
or the most general interference. Turn on the
transmitter and check all channels to determine the extent of the interference and the
number of channels affected. Then disconnect
the antenna and substitute a group of 100 -watt
lamps as a dummy load for the transmitter. Experience has shown that 8 100 -watt lamps connected in two seriesed groups of four in parallel will take the output of a kilowatt transmitter on 28 Mc. if connections are made symmetrically to the group of lamps. Then note
the interference. Now remove plate voltage
from the final amplifier and determine the extent of interference caused by the exciter
stages.
In the average case, when the final amplifier is a beam tetrode stage and the exciter is
.
10' PICKUP WIRE
5T
X0.3
DIA.18E,
0.5 LONG
COVERAGE
-35-140 MC.
IN 34
Figure 6
Crystal -diode wavemeter suitable for checking high-intensity harmonics in TV region.
www.americanradiohistory.com
372
relatively low powered and adequately shielded, it will be found that the interference drops
materially when the antenna is removed and a
dummy load substituted. It will also be found
in such an average case that the interference
will stop when the exciter only is operating.
RADIO
THE
TV and Broadcast Interference
Lx
L4
L3
L5
Lip
l
T
It should be made clear at this
point that the l e v e l of power
used at the transmitter is not of
great significance in the basic harmonic reduction problem. The difference in power level
between a 20 -watt transmitter and one rated
at a kilowatt is only a matter of about 17 db.
Yet the degree of harmonic attenuation required to eliminate interference caused by
harmonic radiation is from 80 to 120 db, depending upon the TV signal strength in the
vicinity. This is not to say that it is not a
simpler job to eliminate harmonic interference
from a low -power transmitter than from a kilowatt equipment. It is simpler to suppress harmonic radiation from a low -power transmitter
simply because it is a much easier problem to
shield a low -power unit, and the filters for the
leads which enter the transmitter enclosure
may be constructed less expensively and
smaller for a low -power unit.
Ca
C3
Cz
Transmitter
Power Level
19 -3
Figure
7
FILTER SCHEMATIC DIAGRAMS
LOW -PASS
The filter illustrated at (A) uses mderived terminating half sections at each
end, with three constant -k mid -sections.
The filter at (B) is essentially the same
except that the center section has been
changed to act as on m- derived section
which can be designed to offer maximum
attenuation to channels 2, 4, 5, or 6 in
accordance with the constants given below. Cutoff frequency is 45 Mc. in all
cases. All coils, except L4 in (B) above,
are wound 1/2 "i.d. with 8 turns per inch.
The (A) Filter
C,,C6 -41.5µµ4d. (40 µµfd. will be found suit-
Low -Pass Filters
able.)
After the transmitter has been shielded, and
all power leads have been filtered in such a
manner that the transmitter shielding has not
been rendered ineffective, the only remaining
available exit for harmonic energy lies in the
antenna transmission line. Hence the main
burden of harmonic attenuation will fall on the
filter installed between the output
of the transmitter and the antenna system.
Experience has shown that the low -pass
filter can best be installed externally to the
main transmitter enclosure, and that the transmission line from the transmitter to the low pass filter should be of the coaxial type.
Hence the majority of low -pass filters are designed for a characteristic impedance of 52
ohms, so that RG -8 /U cable (or RG -58/U for
a small transmitter) may be used between the
output of the transmitter and the antenna transmission line or the antenna tuner.
Transmitting -type low -pass filters for amateur use usually are designed in such a manner as to pass frequencies up to about 30 Mc.
without attenuation. The nominal cutoff frequency of the filters is usually between 38
and 45 Mc., and m- derived sections with maximum attenuation in channel 2 usually are included. Well- designed filters capable of carrying any power level up to one kilowatt are
low -pass
C2, C3,
C4-136 µµfd.
(130 to 140
µµid.
may be
used.)
-0.2
ph; 3 S t. no. 14
L6
L2, L4-0.3 ph; 5 t. no. 12
L3, L4, -0.37 ph; 615 t. no. 12
L,,
The (B) Filter with Mid- Section tuned to Channel
2 (58 Mc..)
C
C6
C2, Ca
C3
-41.5
-136
-87
pfd.
µpfd.
(50
µpfd. fixed
and 75
able in parallel.)
L3, L7-0.2 ph;
L2, L3, L3 , L6
3 Si
L4-0.09 ph;
t. no. 14 55" die. by
-0.3
2
µpfd. vari-
t. no. 14
ph;
5
t. no. 12
'4
" long
The (B) Filter with Mid -Sction tuned to Channel
4 (71 Mc.). All components same except that:
-106
-0.33 ph; 6 t.
L4 -0.05 ph; 1)5 t.
C3
L3, L3
long.
no. 12
no. 14,
3
'8 " dia. by
3/8 "
The (B) Filter with Mid -Section tuned to Channel
5 (81 Mc.). Change the following:
C3
-113 µpfd.
L,, L4-0.34 ph; 6 t. no. 12
L4-0.033 ph; 1 t. no. 14 3/8" dia.
The (B) Filter with Mid -Section tuned to Channel
6 (86 Mc.). All comp
is are essentially
the same except that the theoretical value of
La Is changed to 0.03 ph., and the capacitance
of C3 is changed to 117 µpfd.
www.americanradiohistory.com
HANDBOOK
Low
available commercially from several manufacturers. Alternatively, filters in kit form are
available from several manufacturers at a
somewhat lower price. Effective filters may
be home constructed, if the test equipment is
available and if sufficient care is taken in the
construction of the assembly.
Figures 7, 8 and 9 illustrate
high- performance low-pass
filters which are suitable
for home construction. All are constructed in
slip -cover aluminum boxes (ICA no. 29110)
with dimensions of 17 by 3 by 2% inches. Five
aluminum baffle plates have been installed in
the chassis to make six shielded sections
within the enclosure. Feed-through bushings
between the shielded sections are Johnson no.
135 -55.
Both the (A) and (B) f i t e r types are designed for a nominal cut -off frequency of 45
Mc., with a frequency of maximum rejection
at about 57 Mc. as established by the terminating half- sections at each end. Characteristic impedance is 52 ohms in all cases. The
alternative filter designs diagrammed in figure
7B have provision for an additional rejection
trap in the center of the filter unit which may
be designed to offer maximum r ejection in
channel 2, 4, 5, or 6, depending upon which
channel is likely to be received in the area in
question. The only components which must be
changed when changing the frequency of the
maximum rejection notch in the center of the
filter unit are inductors
L4, and L,, and
capacitor C,. A trimmer capacitor has been included as a portion of C, so that the frequency
of maximum rejection can be tuned accurately
to the desired value. Reference to figures 5
and 6 will show the amateur bands which are
Pass
Filters
373
.I
Construction of
Low -Pass Filters
1
L
Figure 8
PHOTOGRAPH OF THE (B) FILTER WITH
THE COVER IN PLACE
most likely to cause interference to specific
TV channels.
Either high -power or low -power components
may be used in the filters diagrammed in figure
7. With the small Centralab TCZ zero- coefficient ceramic capacitors used in the filter
units of figure 7A or figure 7B, power levels
up to 200 watts output may be used without
danger of damage to the capacitors, provided
the filter is feeding a 52 -ohm resistive load.
It may be practicable to use higher levels of
power with this type of ceramic capacitor in
the filter, but at a power level of 200 watts on
the 28 -Mc. band the capacitors run just perceptibly warm to the touch. As a point of interest, it is the current rating which is of significance to the capacitors used in filter s
such as illustrated. Since current ratings for
small capacitors such as these are not readily
available, it is not possible to establish an
accurate power rating for such a unit. The
high -power unit illustrated in figure 9, which
uses Centralab type 850S and 854S capacitors,
Figure 9
PHOTOGRAPH OF THE (B) FILTER WITH COVER REMOVED
The mid -section in this filter is adjusted for maximum rejection of channel 4. Note that the main
coils of the filter are mounted at on angle of about 45 degrees so that there will be minimum
inductive coupling from one section to the next through the holes in the aluminum partitions.
Mounting the coils in this manner was found to give a measurable improvement in the attenuation characteristics of the filter.
www.americanradiohistory.com
374
THE
TV and Broadcast Interference
has proven quite suitable for power levels up
to one kilowatt.
Capacitors
C2, C4, and C, can be standard manufactured units with normal 5 per cent
tolerance. The coils for the end sections can
be wound to the dimensions given (Lt, L6, and
L,). Then the resonant frequency of the series
resonant end sections should be checked with
a grid -dip meter, after the adjacent input or
output terminal has been shorted with a very
short lead. The coils should be squeezed or
spread until resonance occurs at 57 Mc.
The intermediate m- derived section in the
filter of figure 7B may also be checked with a
grid -dip meter for resonance at the correct rejection frequency, after the hot end of L4 has
been temporarily grounded with a low- inductance lead. The variable capacitor portion of
C, can be tuned until resonance at the correct
frequency has been obtained. Note that there
is so little difference between the constants
of this intermediate section for channels 5 and
6 that variation in the setting of C, will tune
to either channel without materially changing
the operation of the filter.
The coils in the intermediate sections of
the filter (Lr,
L4, and L3 in figure 7A, and
L2, L3, L4, and L6 in figure 7B) may be checked
most conveniently outside the filter unit with
the aid of a small ceramic capacitor of known
value and a grid -dip meter. The ceramic capacitor is paralleled a c r o s s the small coil
with the shortest possible leads. Then the assembly is placed atop a cardboard box and the
RADIO
C
L
resonant frequency checked with a grid -dip
meter. A Shure reactance slide rule may be
used to ascertain the correct resonant frequency for the desired L -C combination and the
coil altered until the desired resonant frequency is attained. The coil may then be installed
in the filter unit, making sure that it is not
squeezed or compressed as it is being installed. However, if the coils are wound exactly as given under figure 10, the filter may
be assembled with reasonable assurance that
it will operate as designed.
filter
low -pass
con nected in the output transmission line of the transmitter is capable of affording an enormous degree of harmonic attenuation. However, the
filter must be operated in the correct manner
or the results obtained will not be up to expectations.
In the first place, all direct radiation from
the transmitter and its control and power leads
must be suppressed. This subject has been
discussed in the previous section. Secondly,
the filter must be operated into a load impedance approximately equal to its design characteristic impedance. The filter itself will
Using Low -Pass
Filters
The
Figure 10
SCHEMATIC OF THE SINGLE -SECTION
HALF -WAVE FILTER
The constants given below are for a characteristic impedance of 52 ohms, for use with
RG -8 /U and RG -58/1/ cable. Coil Lt should
be checked for resonance at the operating
frequency with Cr, and the same with L2 and
C4. This check can be mode by soldering a
low- inductance grounding strap to the lead
between L1 and L2 where it passes through
the shield. When the coils have been trimmed
to resonance with a grid -dip meter, the
grounding strap should of course be removed.
This filter type will give an attenuation of
about 30 db to the second harmonic, about
48 db to the third, about 60 db to the fourth,
fifth, and so on increasing at a
rate of about 30 db per octave.
Ct,C2,C3,C4- Silver mica or small ceramic for low
power, transmitting type ceramic for high power.
Capacitance for different bands is given below:
160 meters -1700 µµfd.
67 to the
-850
-440
-220
meters -110
80 meters
40 meters
20 meters
µµfd.
µµfd.
µµid.
µµfd.
µµfd.
10
6 meters
be made up of
-60
L, -May
sections of B&W Mini ductor for power levels below 250 watts, or of
12
to one kilowatt. Apfor
power
up
no.
enom.
proximate dimensions for the coils are given
below, but the coils should be trimmed to resonate at the proper frequency with a grid -dip meter as discussed above. All coils except the
ones for 160 meters are wound 8 turns per Inch.
160 meters -4.2 µh; 22 turns no. 16 enam.,
dia. 2" long
80 meters -2.1 µh; 13 t. 1" dia. (No. 3014 Mini ductor or no. 12)
40 meters -1.1 µh; 8 t. 1" dla. (No. 3014 or no.
L
1
12 at 8 t.p.i.)
20 meters -0.55 µh; 7
no. 12 at 8 t.p.i.)
t',l"
dia. (No. 3010 or
meters -0.3 µh; 6 t. S4" dia. (No. 3002 or
no. 12 of 8
6 meters -0.17 µh; 4 t. 14" dia. (No. 3002 or
no. 12 at 8 t.p.i.)
10
tpi.)
have very low losses (usually less than 0.5
db) when operated into its nominal value of resistive load. But if the filter is mis- terminated
its losses will become excessive, and it will
not present the correct value of load impedance to the transmitter.
If a f i l t e r, being fed from a high -power
transmitter, is operated into an incorrect termination it may be damaged; the coils may be
overheated and the capacitors destroyed as a
result of excessive r -f currents. Hence it is
wise, when first installing a low -pass filter,
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HANDBOOK
Figure
Broadcast
Interference
375
11
HALF -WAVE FILTER
FOR THE 28 -MC. BAND
Showing one possible type
of construction of o 52 -ohm
half -wave filter for relatively low power operation on
the 28 -Mc. bond.
to check the standing -wave ratio of the load
being presented to the output of the filter with
a standing -wave meter of any of the conventional types. Then the antenna termination or
the antenna coupled should be adjusted, with
low power on the transmitter, until the s.w.r.
of the load being presented to the filter is
less than 2.0, and preferably below 1.5.
Half -wave filters ( "Harmonikers") have been discussed in various publications including the
Nov. -Dec. 1949 GE Ham News. Such filters
are relatively simple and offer the advantage
that they present the same value of impedance
at their input terminals as appears as load
across their output terminals. Such filters normally are used as one-band affairs, and they
offer high attenuation only to the third and
higher harmonics. Design data on the half wave filter is given in figure 10. Construction
of half-wave filters is illustrated in figure 11.
Half -Wove Filters
19 -4
Broadcast
Interference
Interference to the reception of signals in
the broadcast band (540 to 1600 kc.) or in the
FM broadcast band (88 to 108 Mc.) by amateur
transmissions is a serious matter to those
amateurs living in densely populated areas.
Although broadcast interference has recently
been overshadowed by the seriousness of television interference, the condition of BCI is
still present.
In general, signals from a transmitter operating properly are not picked up by receivers
tuned to other frequencies unless the receiver
is of inferior design, or is in poor condition.
Therefore, if the receiver is of good design
and is in good repair, the burden of rectifying
the trouble rests with the owner of the interfering station. Phone and c -w stations both
are capable of causing broadcast interference,
key -click annoyance from the code transmitters
being particularly objectionable.
A knowledge of each of the several types
of broadcast interference, their cause, and
methods of eliminating them is necessary for
the successful disposition of this trouble. An
effective method of combating one variety of
interference is often of no value whatever in
the correction of another type. Broadcast interference seldom can be cured by "rule of
thumb" procedure.
Broadcast interference, as covered in this
section refers primarily to standard (amplitude
modulated, 550 -1600 kc.) broadcast. Interference with FM broadcast reception is much
less common, due to the wide separation in
frequency between the FM broadcast band and
the more popular amateur bands, and due also
to the limiting action which exists in all types
of FM receivers. Occasional interference with
FM broadcast by a harmonic of an amateur
transmitter has been reported; if this condition is encountered, it may be eliminated by
the procedures discussed in the first portion
of this chapter under Television Interference.
The use of frequency -modulation transmission by an amateur station is likely to result
in much less interference to broadcast reception than either amplitude-modulated telephony
or straight keyed c.w. This is true because,
insofar as the broadcast receiver is concerned,
the amateur FM transmission will consist of a
plain unmodulated carrier. There will be no
key clicks or voice reception picked up by
the b-c-1 set (unless it happens to be an FM
receiver which might pick up a harmonic of
the signal), although there might be a slight
click when the transmitter is put on or taken
www.americanradiohistory.com
376
THE
TV and Broadcast Interference
Figure
RADIO
13
HIGH -ATTENUATION WAVE -TRAP
CIRCUIT
circuits
may be tuned to the same
frequency for highest attenuation of a strong
signal, or the two traps may be tuned separately for different bands of operation.
The two
Figure
12
WAVE -TRAP CIRCUITS
The circuit at (A) is the most common arrangement, but the circuit at (B) may give
improved results under certain conditions.
Manufactured wave traps for the desired band
of operation may be purchased or the traps
may be assembled from the data given in
figure
14.
off the air. This is one reason why narrow band FM has become so popular with phone
enthusiasts who reside in densely populated
areas.
Interference
Depending upon whether it is
traceable directly to causes
within the station or within
the receiver, broadcast interference may be
divided into two main classes. For example,
that type of interference due to transmitter
over -modulation is at once listed as b e i n g
caused by improper operation, while an interfering signal that tunes in and out with a
broadcast station is probably an indication of
cross modulation or image response in the receiver, and the poorly- designed input stage
of the receiver is held liable. The various
types of interference and recommended cures
will be discussed in the following paragraphs.
Classifications
This is not a tunable effect, but
a total blocking of the receiver.
A more or less complete "washout" covers
the entire receiver range when the carrier is
switched on. This produces either a complete
blotting out of all broadcast stations, or else
knocks down their volume several decibels depending upon the severity of the interference. Voice modulation of the carrier causing
the blanketing will be highly distorted or even
Blanketing
unintelligible. Keying of the carrier which
produces the blanketing will cause an annoying fluctuation in the volume of the broadcast
signals.
Blanketing generally occurs in the immediate neighborhood (inductive field) of a powerful transmitter, the affected area being directly proportional to the power of the transmitter. Also it is more prevalent with transmitters which operate in the 160 -meter and
80 -meter bands, as compared to those on the
higher frequencies.
The remedies are to (1) shorten the receiving antenna and thereby shift its resonant frequency, or (2) remove it to the interior of the
building, (3) change the direction of either
the receiving or transmitting antenna to minimize their mutual coupling, or (4) keep the
interfering signal from entering the receiver
input circuit by installing a wavetrap tuned
to the signal frequency (see figure 12) or a
low-pass filter as shown in figure 21.
A suitable wave -trap is quite simple in construction, consisting only of a coil and midget
variable capacitor. When the trap circuit is
tuned to the frequency of the interfering signal, little of the interfering voltage reaches
the grid of the first tube. Commercially manufactured wave -traps are available from several
concerns, including the J. W. Miller Co. in
Los Angeles. However, the majority of amateurs prefer to construct the traps from spare
components selected from the "junk box."
The circuit shown in figure 13 is particularly effective because it consists of two
traps. The shunt trap blocks or rejects the
frequency to which it is tuned, while the
series trap across the antenna and ground terminals of the receiver provides a very low impedance path to ground at the frequency to
www.americanradiohistory.com
Wavetraps
HANDBOOK
BAND
COIL,
CAPACITOR,
L
C
1.8 Mc.
Inch no. 30 enom.
75 -,..ofd. var.
1
closewound on 1" form
3.5 Mc.
42 turns no. 30 enam. 50 -cAfd. var.
closewound on 1" form
7.0 Mc.
23
turns no. 24 enam. 50 -µsfd. var.
closewound on 1" form
14
Mc.
10 turns no. 24 enam. 50 -Aµfd. var.
closewound on 1" form
21
Mc.
7
28
Mc.
4
50
Mc.
3
turns no. 24 enam. 50 -ccfd. var.
closewound on 1" form
turns no. 24 enam. 25 -µµfd. vor.
closewound on 1" form
t
spaced
I
no. 24 cram.
on 1" form
/2"
Figure
25 -Acfd.
Figure 15
MODIFICATION OF THE FIGURE
CIRCUIT
var.
circuit arrangement the paralll -tuned
tank is inductively coupled to the antenna
lead with a 3 to 6 turn link instead of being
placed directly in series with the antenna
lead.
which it is tuned and by- passes the signal to
ground. In moderate interference cases, either
the shunt or series trap may be used alone,
while similarly, one trap may be tuned to one
of the frequencies of the interfering transmitter and the other trap to a different interfering frequency. In either case, each trap is
effective over but a small frequency range
and must be readjusted for other frequencies.
The wave -trap must be installed as close
to the receiver antenna terminal as practicable, hence it should be as small in size as
possible. The variable capacitor may be a
midget air -tuned trimmer type, and the coil
may be wound on a 1 -inch dia. form. The table
of figure 14 gives winding data for wave -traps
built around standard variable capacitors. For
best results, both a shunt and a series trap
should be employed as shown.
Figure 15 shows a two- circuit coup e d
wave -trap that is somewhat sharper in tuning
and more efficacious. The specifications for
the secondary coil L, may be obtained from
the table of figure 14. The primary coil of the
shunt trap consists of 3 to 5 closewound turns
of the same size wire wound in the same direction on the same form as L, and separated
1
latter by
Overmodulation
A
'4
13
In this
14
COIL AND CAPACITOR TABLE FOR
AMATEUR -BAND WAVETRAPS
from the
377
of an inch.
carrier modulated in excess
of 100 per cent acquires
sharp cutoff periods which give rise to transients. These transients create a broad signal
and generate spurious responses. Transients
caused by overmodulation of a radio -telephone
signal may at the same time bring about impact or shock excitation of nearby receiving
antennas and power lines, generating interfering signals in that manner.
Broadcast interference due to overmodulation is frequently encountered. The remedy is
to reduce the modulation percentage or to use
a clipper -filter system or a high -level splatter
suppressor in the speech circuit of the transmitter.
Cross modulation or cross talk is
characterized by the amateur signal riding in on top of a strong
broadcast signal. There is usually no heterodyne note, the amateur signal being tuned in
and out with the program carriers.
This effect is due frequently to a faulty input stage in the affected receiver. Modulation
of the interfering carrier will swing the operating point of the input tube. This type of
trouble is seldom experienced when a variable-µ tube is used in the input stage.
Where the receiver is too ancient to incorporate such a tube, and is probably poorly
shielded at the same time, it will be better to
attach a wave -trap of the type shown in figure
12 rather than to attempt rebuilding of the receiver. The addition of a good ground and a
shield can over the input tube often adds to
the effectiveness of the wave -trap.
Cross
Modulation
Transmission via
Capacitive Coupling
A
small
amount of ca-
pacitive coupling is now
widely used in receiver
r.f. and antenna transformers as a gain booster
at the high- frequency end of the tuning range.
The coupling capacitance is obtained by
means of a small loop of wire cemented close
to the grid end of the secondary winding, with
one end directly connected to the plate or antenna end of the primary winding. (See figure
16.)
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378
THE
TV and Broadcast Interference
CAPACITIVE
COUPLING LOOP
Figure 16
CAPACITIVE BOOST COUPLING
CIRCUIT
Such circuits, included within the broadcast
receiver to bring up the stage gain at the
high -frequency end of the tuning ronge, have
a tendency to increase the susceptibility of
the receiver to interference from amateur band transmissions.
It is easily seen that a small capacitor at
this position will favor the coupling of the
higher frequencies. This type of capacitive
coupling in the receiver coils will tend to
pass amateur high- frequency signals into a receiver tuned to broadcast frequencies.
The amount of capacitive coupling may be
reduced to eliminate interference by moving
the coupling turn further away from the secondary coil. However, a simple wave -trap of
the type shown in figure 12, inserted at the
antenna input terminal, will generally accomplish the same result and is more to be recommended than reducing the amount of capacitive coupling (which lowers the receiver gain
at the high- frequency end of the broadcast
band). Should the wave -trap alone not suffice,
it will be necessary to resort to a reduction
in the coupling capacitance.
In some simple broadcast receivers, capacitive coupling is obtained by closely coupled
primary and secondary coils, or as a result of
running a long primary or antenna lead close
to the secondary coil of an unshielded antenna coupler.
strong local carriers applied to a non -linear impedance,
the beat note resulting from cross -modulation
between them may fall on some frequency
within the broadcast band and will be audible
at that point. If such a "phantom" signal falls
on a local broadcast frequency, there will be
heterodyne interference as well. This is a
common occurence with broadcast receivers in
the neighborhood of two amateur stations, or
an amateur and a police station. It also sometimes occurs when only one of the stations is
located in the immediate vicinity.
Phantoms
With two
RADIO
As an example: an amateur signal on 3514
kc. might beat with a local 2414 -kc. police
carrier to produce a 1100 -kc. phantom. If the
two carriers are strong enough in the vicinity
of a circuit which can cause rectification, the
1100 -kc. phantom will be heard in the broadcast band. A poor contact between two oxidized wires can produce rectification.
Two stations must be transmitting simultaneously to produce a phantom signal; when
either station goes off the air the phantom
disappears. Hence, this type of interference
is apt to be reported as highly intermittent and
might be difficult to duplicate unless a test
oscillator is used "on location" to simulate
the missing station. Such interference cannot
be remedied at the transmitter, and often the
rectification takes place some distance from
the receivers. In such occurrences it is most
difficult to locate the source of the trouble.
It will also be apparent that a phantom
might fall on the intermediate frequency of a
simple superhet receiver and cause interference of the untunable variety if the manufacturer has not provided an i -f wave -trap in the
antenna circuit.
This particular type of phantom may, in
addition to causing i-f interference, generate
harmonics which may be tuned in and out with
heterodyne whistles from one end of the receiver dial to the other. It is in this manner
that birdies often result from the operation of
nearby amateur stations.
G
hen one component of a
phantom is
a
steady, unmodulated carrier, only the intelligence present on the other carrier is conveyed
to the broadcast receiver.
Phantom signals almost always may be
identified by the suddenness with which they
are interrupted, signalizing withdrawal of one
party to the union. This is especially baffling
to the inexperienced interference -locater, who
observes that the interference suddenly disappears, even though his own transmitter remains in operation.
If the mixing or rectification is taking place
in the receiver itself, a phantom signal may
be eliminated by removing either one of the
contributing signals from the receiver input
circuit. A wave -trap of the type shown in figure 12, tuned to either signal, will do t h e
trick. If the rectification is taking place outside the receiver, the wave -trap should be
tuned to the frequency of the phantom, instead
of to one of its components. I -f wave -traps
may be built around a 2.5- millihenry r -f choke
as the inductor, and a compression -type mica
padding capacitor. The capacitor should have
a capacitance range of 250 -525 µµfd. for the
175- and 206 -kc. intermediate frequencies;
65 -175 µµfd. for 260 -kc. and other intermedi-
www.americanradiohistory.com
HANDBOOK
Audio
ates lying between 250- and 400 -kc; and
17 -80 µµEd. for 456 -, 465-, 495 -, and 500 -kc.
Slightly more capacitance will be required for
resonance with a 2.1 millihenry choke.
This sort of interference arises
from the transmitter itself. The
radiation of any signal (other than
the intended carrier frequency) by an amateur
station is prohibited by FCC regulations. Spurious radiation may be traced to imperfect neutralization, parasitic oscillations in the r -f or
modulator stage s, or to "broadcast- band"
variable- frequency oscillators or e.c.o.'s.
Low- frequency parasitics may actually occur on broadcast frequencies or their near sub harmonics, causing direct interference to programs. An all -wave monitor operated in the
vicinity of the transmitter will detect these
spurious signals.
The remedy will be obvious in individual
cases. Elsewhere in this book are discussed
methods of complete neutralization and the
suppression of parasitic oscillations in r -f
and audio stages.
Rectification
379
HIGH -MU TUBE
SUCH AS 12507
Spurious
Emissions
Inexpensive tab e- model
a -c /d -c receivers are particularly susceptible to interference from amateur transmissions. In fact, it may be said
with a fair degree of assurance that the majority of BCI encountered by amateurs operating
in the 1.8 -Mc. to 29-Mc. range is a result of
these inexpensive receivers. In most cases
the receivers are at fault; but this does not
absolve the amateur of his responsibility in
attempting to eliminate the interference.
A -c /d -e Receivers
1
cases of interference
receivers, par-
Stray Receiver
In most
Rectification
to inexpensive
ticularly those of the a -c /d -c
type, it will be found that stray receiver rectification is causing the trouble. The offending
stage usually will be found to be a high -mu
triode as the first audio stage following the
second detector. Tubes of this type are quite
non-linear in their grid characteristic, and
hence will readily rectify any r -f signal appearing between grid and cathode. The r -f signal may get to the tube as a result of direct
signal pickup due to the lack of shielding, but
more commonly will be fed to the tube from
the power line as a result of the series heater
string.
The remedy for this condition is simply to
insure that the cathode and grid of the high -mu
audio tube (usually a 12SQ7 or equivalent) are
at the same r-f potential. This is accomplished
by placing an r-f by -pass capacitor with the
shortest possible leads directly from grid to
cathode, and then adding an impedance in the
lead from the volume control to the grid of the
Figure 17
CIRCUITS FOR ELIMINATING AUDIO
-
STAGE RECTIFICATION
audio tube. The impedance may be an amateur
band r -f choke (such as a National R -I00U)
for best results, but for a majority of cases
it will be found that a 47,000 -ohm V2-watt resistor in series with this lead will giv.e satisfactory operation. Suitable circuits for such
an operation on the receiver are given in figure 17.
In many a.c. -d.c. receivers there is no r -f
by -pass Included across the plate supply rectifier for the set. If th e r e is an appreciable
level of r -f signal on the power line feeding
the re ce i v e r, r -f rectification in the power
rectifier of the receiver can cause a particularly bad type of interference which may be
received on other broadcast receivers in the
vicinity in addition to the one causing the
rectification. The soldering of a 0.01 -pfd. disc
ceramic capacitor directly from anode to cathode of the power rectifier (whether it is of the
vacuum -tube or selenium- rectifier type) usually will by -pass the r -f signal across the rectifier and thus eliminate the difficulty.
Several sets have been
encountered where there
was only a slightly interfering signal; but, upon placing one's hand up
to the volume control, the signal would greatly increase. Investigation revealed that the
volume control was installed with its shaft
insulated from ground. The control itself was
connected to a critical part of a circuit, in
many instances to the grid of a high -gain audio stage. The cure is to install a volume control with all the terminals insulated from the
shaft, and then to ground the shaft.
"Floating" Volume
Control Shafts
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380
THE
TV and Broadcast Interference
r-i BO-
RADIO
METAL
BAND
3.5 Mc.
COIL,
L
CAPACITOR,
17 turns
no. 14 enameled 100-;.,.íd.
-inch diameter
23/4-inch length
variable
3
11
7.0 Mc.
C
C
TO A.C.
LINE
turns
I
no. 14 enameled
212 -inch
100- n,.fd.
diameter
length
SHIELD BRAID
LOV
C
-
variable
I
Qv
L
TO TRANSMITTER
OR RECEIVER
I
r
SHIELD BRAID
J
l'2 -inch
14
21
and
Mc.
no.
4 turns
10 enameled 100 -,. ;.td.
3 -inch
diameter
118 -inch length
3
27 and
28 Mc.
turns
I ¡ -inch o.d.
copper
tubing
2 -inch
1
variable
diameter
-inch length
100- ,,id.
variable
Figure 19
RESONANT POWER -LINE
WAVE -TRAP CIRCUIT
The resonant type of power -line filter is
more effective than the more conventional
"brute force type of line filter, but requires
tuning to the operating frequency of the
transmitter.
Figure 18
COIL AND CAPACITOR TABLE
FOR A -C LINE TRAPS
a hen radio - frequency energy
from a radio transmitter enters a
broadcast receiver through the
a -c power lines, it has either been fed back
into the lighting system by the offending transmitter, or picked up from the air by over -head
power lines. Underground lines are seldom rePower -Line
Pickup
sponsible for spreading this interference.
To check the path whereby the interfering
signals reach the line, it is only necessary to
replace the transmitting antenna with a dummy antenna and adjust the transmitter for maximum output. If the interference then ceases,
overhead lines have been picking up the energy. The trouble can be cleared up by installing a wave-trap or a commercial line filter in
the power lines at the receiver. If the receiver
is reasonably close to the transmitter, it is
very doubtful that changing the direction of
the transmitting antenna to right angles with
the overhead lines will eliminate the trouble.
If, on the contrary, the interference continues when the transmitter is connected to
the dummy antenna, radio- frequency energy is
being fed directly into the power line by the
transmitter, and the station must be inspected
to determine the cause.
One of the following reasons for the trouble
will usually be found: (1) the r -f stages are
not sufficiently bypassed and /or choked, (2)
the antenna coupling system is not performing
efficiently, (3) the power transformers have
no electrostatic shields; or, if shields are present, they are ungrounded, (4) power lines are
running too close to an antenna or r -f circuits
carrying high currents. If none of these causes
apply, wave -traps must be installed in the
power lines at the transmitter to remove r-f
energy passing back into the lighting system.
The wave -traps used in the power lines at
transmitter or receiver must be capable of
passing relatively high current. The coils are
accordingly wound with heavy wire. Figure 18
lists the specifications for power line wave trap coils, while figure 19 illustrates the method of connecting these wave- traps. Observe
that these traps are enclosed in a shield box
of heavy iron or steel, well grounded.
All -Wave
Each complete- coverage home receiver is a potential source of annoyance to the transmitting amateur. The novice short -wave broadcast- listener
who tunes in an amateur station often considers it an interfering signal, and complains
Receivers
accordingly.
Neither selectivity nor image rejection in
most of these sets is comparable to t ho s e
properties in a communication receiver. The
result is that an amateur signal will occupy
too much dial space and appear at more than
one point, giving rise to interference on adjacent channels and distant channels as well.
If carrier- frequency harmonics are present
in the amateur transmission, serious interference will result at the all -wave receiver. The
harmonics may, if the carrier frequency has
been so unfortunately chosen, fall directly
upon a favorite short -wave broadcast station
and arouse warranted objection.
The amateur is apt to be blamed, too, for
transmissions for which he is not responsible,
so great is the public ignorance of short -wave
allocations and signals. Owners of all -wave
receivers have been quick to ascribe to amateur stations all signals they hear from tape
machines and V- wheels, as well as stray tones
and heterodyne
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flutters.
HANDBOOK
The amateur cannot be held responsible
when his carrier is deliberately tuned in on
all -wave receiver. Neither is he accountable for the width of his signal on the receiver
dial, or for the strength of image repeat points,
if it can be proven that the receiver design
does not afford good selectivity and image rean
jection.
If he so desires, the amateur (or the owner
of the receiver) might sharpen up the received
signal somewhat by shortening the receiving
antenna. Set retailers often supply quite a
sizeable antenna with all -wave receivers, but
most of the time these sets perform almost as
well with a few feet of inside antenna.
The amateur is accountable for harmonics
of his carrier frequency. Such emissions are
unlawful in the first place, and he must take
all steps necessary to their suppression. Practical suggestions for the elimination of harmonics have been given earlier in this chapter under Television Interference.
Image Interference
Interference
Image
In addition to those types
of interference al read y
discussed, there are two more which are common to superhet receivers. The prevalence of
these types is of great concern to the amateur, although the responsibility for their existence more properly rests with the broadcast
receiver.
The mechanism whereby image production
takes place may be explained in the following
manner: when the first detector is set to the
frequency of an incoming signal, the high -frequency oscillator is operating on another frequency which differs from the signal by the
number of kilocycles of the intermediate frequency. Now, with the setting of these two
stages undisturbed, there is another signal
which will beat with the high- frequency oscillator to produce an i -f signal. This other signal is the so- called image, which is separated
from the desired signal by twice the intermediate frequency.
Thus, in a receiver with 175 -kc. i.f., tuned
to 1000 kc.: the h -f oscillator is operating on
1175 kc., and a signal on 1350 kc. (1000 kc.
plus 2 x 175 kc.) will beat with this 1175 kc.
oscillator frequency to produce the 175 -kc. i -f
signal. Similarly, when the same receiver is
tuned to 1400 kc., an amateur signal on 1750
kc. can come through.
If the image appears only a few cycles or
kilocycles from a broadcast carrier, heterodyne
interference will be present as well. Otherwise, it will be tuned in and out in the manner
of a station operating in the broadcast band.
Sharpness of tuning will be comparable to that
of broadcast stations producing the same a -v -c
voltage at the receiver.
381
The second variety of superhet interference
is the result of harmonics of the receiver h -f
oscillator beating with amateur carriers to produce the intermediate frequency of the receiver. The amateur transmitter will always be
found to be on a frequency equal to some harmonic of the receiver h -f oscillator, plus or
minus the intermediate frequency.
As an example: when a broadcast superhet
with 465 -kc. i.f. is tuned to 1000 kc., its high frequency oscillator operates on 1465 kc. The
third harmonic of this oscillator frequency is
4395 kc., which will beat with an amateur signal on 3930 kc. to send a signal through the
i -f amplifier. The 3930 kc. signal would be
tuned in at the 1000 -kc. point on the dial.
Some oscillator harmonics are so related to
amateur frequencies that more than one point
of interference will occur on the receiver dial.
Thus, a 3500 -kc. signal may be tuned in at
six points on the dial of a nearby broadcast
superhet having 175 kc. i.f. and no r -f stage.
Insofar as remedies for image and harmonic
superhet interference are concerned, it is well
to remember that if the amateur signal did not
in the first place reach the input stage of the
receiver, the annoyance would not have been
created. It is therefore good policy to try to
eliminate it by means of a wave-trap or low pass filter. Broadcast superhets are not always the acme of good shielding, however,
and the amateur signal is apt to enter the circuit through channels other than the input circuit. If a wave-trap or filter will not cure the
trouble, the only alternative will be to attempt
OUTPUT
INPUT
T
O
CONSTANT
K
INPUT
O
O
TYPE
FREQUENCY
OUTPUT
T
T
M- DERIVED TYPE
FREQUENCY
Figure
20
TYPES OF LOW -PASS FILTERS
Filters such as these may be used in the
circuit between the antenna and the input
of the receiver.
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382
TV and Broadcast Interference
L,
ANT
L2
TO
TC2
T`
Cs
RECEIVER ANT. POST
TCA
1
O TO RECEivER CND. POST
GND.
Figure
21
COMPOSITE LOW-PASS FILTER
CIRCUIT
This filter is highly effective in reducing
broadcast interference from all high frequency stations, and requires no tuning. Constants for 400 ohm terminal impedance and
1600 kc. cutoff are as follows: L,, 65 turns
no. 22 d.c.c. closewound on 1%2 in. dia. form.
L2, 41 turns ditto, not coupled to Lr. C,,
250 µµfd. fixed mica capacitor. C2, 400 Auld.
fixed mica capacitor. C3 and C4, ISO Auld.
fixed mica capacitors, former of S% tolerance. With some receivers, better results
will be obtained with o 200 ohm carbon resistor inserted between the filter and antenna post on the receiver. With other receivers the effectiveness will be improved
with a 600 ohm carbon resistor placed from
the antenna post to the ground post on the
receiver. The filter should be placed as
close to the receiver terminals os possible.
to select a transmitter frequency such that
neither image nor harmonic interference will be
set up on favorite stations in the susceptible
receivers. The equation given earlier may be
used to determine the proper frequencies.
Pass Filters
The greatest drawback of
the wave -trap is the fact
that it is a single- frequency device; i.e. -it
may be set to reject at one time only one frequency (or, at best, an extremely narrow band
of frequencies). Each time the frequency of
the interfering transmitter is changed, every
wave -trap tuned to it must be retuned. A much
more satisfactory device is the wave filler
which requires no tuning. One type, the low pass filter, passes all frequencies below one
critical frequency, and eliminates all higher
frequencies. It is this property that makes the
device ideal for the task of removing amateur
Low
frequencies from broadcast receivers.
A good low -pass filter designed for maximum attenuation around 1700 kc. will pass
all broadcast carriers, but will reject signals
originating in any amateur band. Naturally
such a device should be installed only in
standard broadcast receivers, never in all wave sets.
Two types of low -pass filter sections are
shown in figure 20. A composite arrangement
comprising a section of each type is more
effective than either type operating alone. A
composite filter composed of one K- section
and one shunt -derived M- section is shown in
figure 21, and is highly recommended. The
M- section is designed to have maximum attenuation at 1700 kc., and for that reason C,
should be of the "close tolerance" variety.
Likewise, C, should not be stuffed down inside L, in the interest of compactness, as
this will alter the inductance of the coil appreciably, and likewise the resonant frequency.
If a fixed 150 µµfd. mica capacitor of 5 per
cent tolerance is not available for C1r a compression trimmer covering the range of 125175 µµEd. may be substituted and adjusted to
give maximum attenuation at about 1700 kc.
19 -5
HI -FI Interference
The rapid growth of high -fidelity sound systems
the home has brought about many cases of
interference from a nearby amateur transmitter.
In most cases, the interference is caused by
stray pickup of the r-f signal by the interconnecting leads of the hi -fi system and audio rectification in the.low level stages of the amplifier.
The solution to this difficulty, in general, is to
bypass and filter all speaker and power leads
to the hi -fi amplifier and preamplifier. A combination of a VHF choke and 500 µpfd ceramic
disc capacitors in each power and speaker lead
will eliminate r -f pickup in the high level section
of the amplifier. A filter such as shown in figure
17A placed in the input circuit of the first audio
stage of the preamplifier will reduce the level of
the r -f signal reaching the input circuit of the
amplifier. To prevent loss of the higher audio
frequencies it may be necessary to decrease the
value of the grid bypass capacitor to 50 ppfd
or so.
Shielded leads should be employed between the
amplifier and the turntable or f -m tuner. The
shield should be. grounded at both ends of the
line to the chassis of the equipment, and care
should be taken to see that the line does not
approach an electrical half -wavelength of the
radio signal causing the interference. In some
instances, shielding the power cable to the hi -fi
equipment will aid in reducing interference. The
framework of the phonograph turntable should be
grounded to the chassis of the amplifier to reduce
stray r -f pickup in the turntable equipment.
in
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562
THE RADIO
Receivers and Transceivers
(figure 37). The front panel has the same
dimensions as the outside of this box, and
takes the form of a shallow pan, about 3/4 -inch
deep (figure 36). The panel is affixed to two
angle brackets mounted on the edges of the
sub -panel. The two meters are mounted to the
sub -panel, as is the dial mechanism and the
pilot lamp. The various potentiometers are
mounted to small L- shaped plates spaced away
from the sub -panel. The pan -shaped panel is
merely a decorative cover that finishes the
appearance of the unit.
The dial is home -made and is driven by a
35 -1 gear train made from re- mounted parts
of a surplus BC -453 ( "Command ") receiver
dial (figure 37). The dial drive and pointer
may be made from a broadcast -type slide rule
dial and the escutcheon is cut and formed
from a piece of bakelite and is suitably
engraved.
Figure 38
UNDER -CHASSIS VIEW OF TRANSCEIVER
Neat wiring and use of cabling techniques makes "clean" looking assembly. Power leads and
long "runs" are laced into main cable passing in a square about r.f. section. Small components are
soldered directly to socket pins, or are mounted on phenolic terminal boards, as is the case of the
squelch components. R.I. coils and padding capacitors are at center of layout, beneath main
tuning capacitor gang. Individual shield sections separate the r.f. stages. Change -over relay RY
is mounted to rear wall of chassis next to antenna receptacle.
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HANDBOOK
Deluxe Mobile Transceiver
561
Figure 37
VIEW OF TRANSCEIVER WITH FRONT PANEL REMOVED
The various panel controls are mounted on L- shaped brackets attached to the sub -panel by means
of metal bushings. Meters are mounted to the sub -panel by means of encircling straps. The dial
mechanism is made from geared portions of "command" receiver dial drive fixed to a thin
phenolic plate.
switches the 250 -volt supply from the receiver
section to the transmitter section and section B
transfers the antenna from the receiver to the
transmitter. It is necessary to remove the B -plus
from the modulator and power amplifier of
the transmitter during reception, and this may
be accomplished by switching off the high
voltage supply by means of an auxiliary relay
whose actuator coil is paralleled with the coil
of relay RY1. The auxiliary relay should be
located at the power supply.
Transceiver
Construction
This transceiver is an excellent
example of the fine workmanship possible by an amateur
adept in sheet metal work and who has the
necessary shop facilities. The chassis -cabinet is
made of 14 -gauge sheet durai, cut and bent to
size by a sheet metal shop. The assembly is
made up of six pieces: A wrap- around back
and side piece, removable top and bottom
plates, the chassis, the sub -panel, and the front
panel. Ventilation holes are drilled in the top
plate and the wrap -around section to ventilate the unit, as a considerable amount of heat
is generated by the tubes.
The chassis is constructed with a 1/2 -inch
lip around the edges which is bolted to the
wrap- around piece and the sub -panel. In order
to conserve height, the chassis has a "step" in
it to allow room for the taller tubes (6146
and 6BQ6 -GT's) and the modulation transformer. Less room above the chassis is required for the receiver section, and a correspondingly greater area beneath the chassis
allows room for the receiver coils and stage
shields. The "step" can be seen in figure 36,
running from the front to the back of the
chassis, immediately to the right of the ganged
tuning capacitors.
The chassis, the wrap- around piece, and the
sub -panel make up a complete TVI -proof box
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560
THE RADIO
Receivers and Transceivers
transformer coupled to two 6BQ6 -GT pentodes connected as zero bias class B modulators. No grid bias or screen voltage is
required for the modulator, and the audio
driving voltage is applied to the screens of
the tubes. The control grid is connected to
PLACEMENT OF MAJOR COMPONENTS
the cathode. This simple circuit is capable of
over 40 watts of audio output. Negative peak
control is exercised by a silicon rectifier placed
in series with the secondary winding of the
modulation transformer, and a simple low
pass audio filter composed of the leakage
reactance of the modulation transformer plus
the plate bypass capacitor of the r.f. amplifier
stage reduces the higher order audio harmonics
generated by this system. A high level of "talk
power" is thus insured.
The receiver section of the unit occupies the
chassis, with the
transmitter section at right. The "step" in the
chassis is at the right of the main tuning
Filament and
Control Circuits
Figure 36
TOP VIEW OF TRANSCEIVER SHOWS
left -hand section of the
gang, running parallel to it, from the front to
the rear of the chassis.
Receiver i.f. section runs along left edge of
chassis, with squelch, regulator tube, and
second conversion oscillator in the adjacent
row. R.I. and audio stages are next to tuning
gang. The three sections of capacitors nearest
the front panel are for the receiver portion,
while the rear capacitors are for the transmitter section. The 6146 plate tank coil is at
the rear of the chassis.
Modulator section occupies right -hand portion
of chassis, with transmitter r.f. stages immediately to the left. 6CL6 buffer tube is
shielded and directly in front of the 6146.
The transmitter is designed
for either 6- or 12 -volt
operation. The circuit of
figures 34 -35 shows 6 -volt configuration. For
12 -volt operation, it is only necessary to rewire the power plugs as shown and filament
switching is automatic.
Change -over from receive to transmit is
accomplished by means of relay RY1, which is
actuated by the microphone button, and which
has a d.c. coil suited to the voltage of the automobile battery. Contact section A of this relay
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HANDBOOK
Deluxe Mobile Transceiver
stability. The oscillator runs continuously and
is voltage regulated.
A single i.f. transformer (T1) provides
sufficient image selectivity at 4.26 Mc. and
no additional amplification or tuned circuits
are required. The second intermediate frequency is 260 kc., and a 6BE6 multi -grid
converter tube is used as a mixer to this frequency. The local oscillator is crystal controlled at 4.52 Mc., and makes use of the
6BE6 as a "hot cathode' crystal oscillator.
Precise adjustment of the oscillator frequency
may be made by means of the variable inductance (L;) in the grid- cathode circuit of the
mixer tube. The choice of frequency of the
mixing oscillator is important in that no harmonic frequencies of the oscillator should fall
into the 10 meter band, or into its "image"
frequency band. This insures that undesired
"birdies" or spurious responses of the receiver
are reduced to an absolute minimum.
Two stages of i.f. amplification employing low
filament drain 6BJ6 tubes provide sufficient receiver gain, and are followed by a 6AL5 detector/a.v.c. rectifier stage. A two -stage noise
limiter patterned after the popular "twin noise squelch (TNS) circuit" provides maximum noise rejection with minimum audio
distortion. A 12AX7 and 6AL5 are used in
this portion of the receiver. A 12AT7 tube
serves a dual purpose as a first audio stage
and v.t.v.m. -type S -meter amplifier, followed
by a 6ÁQ5 audio output stage. The S -meter
circuit makes use of a "backwards reading"
meter that rests at full scale. The a.v.c. voltage
applied to the amplifier tube reduces the meter
current in accordance with the strength of the
incoming signal.
The Transmitter Section. The transmitter section of the transceiver is shown in figure 35,
and in outline form in figure 32. A 12AT7
dual triode serves as a mixer -oscillator stage,
beating the receiver v.f.o. with a 4.26 Mc.
crystal (equal to the receiver intermediate frequency) The sum of these two frequencies is
the transmitting frequency, which is equal to
the frequency of reception. Following the
mixer -oscillator are two gang -tuned r.f. amplifier stages employing high gain 6CL6 pentode tubes. The second stage is neutralized for
maximum stability. The power amplifier stage
uses a single 6146 in a pi- network output
circuit, which is also gang -tuned in conjunction with the exciter and v.f.o.
Tuning and loading controls of the power
amplifier stage are located on the rear of
the chassis and need not be readjusted unless
a change is made in the antenna system (figure
39) Antenna change -over is controlled by a
section of relay RYI. Grid and plate currents
of the 6146 are monitored by meter M2.
The Modulator Section. The modulator is designed to work with either a ceramic -type
crystal microphone, or a high impedance dynamic unit. A 12AT7 serves as a two stage
resistance coupled amplifier, exciting a parallel
connected 12ÁU7 driver. This, in turn, is
.
.
Figure 33
MINIATURE
POWERHOUSE
PACKS PLENTY
OF PUNCH!
The transceiver is
in a custom -made
557
built
case
which permits maximum
utilization of available
space. Mounting flanges
may be seen attached to
upper portion of transceiver case. At left of
main tuning dial are volume control (with on -off
switch), r.f. gain control,
and squelch. At right are
microphone level control,
meter switch, and microphone receptacle.
riwgr-
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THE RADIO
Receivers and Transceivers
556
ANT
R.F. AMP.
1ST MIX
(26 Mc.)
(4.26 AK.)
®
®
2NDI%.
(280
I.F.
I.F
DET.
AVC
RC.
®
®
®
®
GAIN
AUDIO
AUDIO
5-METER
12AT7
®'I
SPMR
(4.26MC)
SPEECH AMP.
DRIVER
J
MIC
Figure 32
BLOCK DIAGRAM OF THE TRANSCEIVER
The tuning oscillator of the unit covers the range of 23.74 -25.44 megacycles. The transmitter
conversion crystal (4.26 Mc.) is the same frequency as the first i.f. of the receiver, thus placing
receiver and transmitter operating frequencies at the same soot on the tuning dial. Receiver
selectivity is obtained by use of two i.f. stages at 260 kc. R.I. circuits of both transceiver sections
are ganged for single dial control.
Circuit
A block diagram of the trans -
ceiver is shown in figure 32.
The circuit utilizes a double
conversion receiver employing eleven tubes
and a voltage regulator, and a v.f.o.- controlled
amplitude modulated transmitter having eight
tubes. A feature of the unit is that transmitting
and receiving frequencies are locked together
and controlled by one master oscillator. All
variable r.f. circuits are tracked for single control tuning. The operator merely tunes the
transceiver to the station he desires to contact,
pushes the microphone control button and the
transmitter is tuned to the same frequency,
ready to "talk."
The Receiver Section. The receiver portion of
the transceiver is shown in figure 34, and in
outline form in figure 32. Double conversion
is used, with the second conversion oscillator
crystal controlled. The first conversion oscillator is also the v.f.o. for the transmitter section, as explained later. The three r.f. circuits
Description
of the receiver section (r.f. stage, mixer, and
oscillator) are gang -tuned for proper tracking
across the 10 meter band.
The r.f. stage utilizes a 6BZ6 high gain,
semi- remote cutoff pentode to achieve maximum signal gain without troublesome cross modulation effects from strong nearby signals.
The circuit of this stage is conventional, except
that the cathode return may be removed from
the gain buss by switch S) for optimum weak
signal response, if desired. Partial a.v.c. is applied to the 6BZ6 by means of a high impedance voltage divider in the a.v.c. system.
A 6BA7 multi -grid converter tube is used
as a mixer from the operating frequency to
the first intermediate frequency of 4.26 Mc.
Mixer injection voltage is applied to the #1
grid of the 6BA7. The local oscillator employs a 6ÁH6 and tunes the range of 23,74025,400 kc., with a slight overlap at both ends
of the range. A high -C "hot cathode" oscillator
circuit is employed for maximum frequency
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HANDBOOK
Deluxe Mobile Transceiver
shown in figure 30. This unit is sufficient
to run one converter at a time.
27 -6
560
10N
8+
70 V.
B+RCG
A Deluxe
Mobile Transceiver
The modern automobile leaves little room
for radio equipment mounted in proximity
to the driver. Mobile equipment, as a result,
must be built more compactly in order to fit
in the dashboard firewall area available for
auxiliary equipment. The amateur having sheet
555
6.3
Ti
=
V.
125 V., 50 MA.
6.3v.,2A
STANCOR
PI-6121
SR= SELENIUM RECTI FIER.
S0 MA.
GND
Figure 30
SCHEMATIC, CONVERTER POWER
SUPPLY
Figure 31
COMPACT TRANSCEIVER OFFERS
ULTIMATE IN MOBILE
COMMUNICATION
This compact a.m. transceiver is a complete
meter station, packaged so that it will fit
into all but the most cramped automobiles.
The transmitter section runs up to 70 watts
input and is designed for "on frequency"
operation with the receiver section. The easy to -read dial controls the master oscillator for
both transmission and reception. The operator
merely tunes the transceiver to the station he
desires to contact and the transmitter is
automatically tuned to the correct frequency.
The transceiver is mounted in the car by
means of dashboard clamps fastened to the
top of the unit by means of a sliding fixture.
Top and bottom plates are removable by
means of snap fasteners, and are perforated
for good ventilation. Simplicity of operation
permits transceiver to be operated without the
driver taking his eyes from the road.
10
metal working facilities at hand is indeed
fortunate, as he may custom-form his equipment chassis and cabinet to fit the space provided in his particular automobile.
Described in this section is a deluxe transceiver, designed and built by W7JNC which
will fit easily into all but the most cramped
automobiles. The unit is a complete 10 meter
station capable of running up to 70 watts
input, having a sensitive double conversion
receiver, and packaged in a cabinet measuring
only 11 inches wide, 4 inches high, and 8
inches deep. The transceiver is suited for
either mobile or fixed -station operation.
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554
Receivers and Transceivers
THE RADIO
Figure 28
PLACEMENT OF MAJOR COMPONENTS ABOVE THE CHASSIS
Figure 29
CLOSE-UP OF
2 -METER R.F.
AMPLIFIER STAGE
A shield partition passes
across the center of Nuvistor socket. The grid
compartment is at the
right, and plate cornpartment at the left. Coil L,
is wound on high value
composition resistor. Six meter r.f. section is identical except for coil
changes.
I,
1
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552
Receivers and Transceivers
verter. Adjust the converter output coil
(L6 /L16) for maximum receiver noise, making
sure that you are not tuning to the image frequency of the receiver. Connection to the
receiver should be made by means of a short
length of coaxial line to prevent spurious
signal pick -up in the 28 -30 Mc. range.
With these preliminary adjustments made,
the r.f. stage is ready for test and alignment.
Start with the 144 Mc. section. Remove the
B+ to coil L3 and insert a 6CW4 in the r.f.
socket. Connect a temporary antenna to the
converter and tune in a strong local test
signal. Make sure signal pickup is via the antenna and not by indirect pickup via coils L3
or L5. Roughly peak coils L3, L ;, and L6 for
Figure 26
UNDER -CHASSIS
VIEW OF "SIAMESE"
CONVERTER
The converter chassis has been removed from
the end plates for this photograph. The two
crystal oscillators are at the center of the
chassis, with the mixer stages adjacent to
them. At the ends of the chassis are the r.f.
amplifiers. Note that a T- shaped shield iso-
lates the input and output circuits of the r.f.
amplifier from the remainder of the circuitry.
The shields are made up of thin flashing
copper and are about 112 inches high. The
small leg of the shield passes across the center
of the Nuvistor socket, and the grid -plate
blocking capacitor passes through a hole
drilled in this partition.
maximum signal. Now, carefully spread and
adjust the turns of coil L2 for minimum received signal. The neutralization point will be
a sharp and almost complete signal null. If
neutralization is obscure, add or remove a
turn or two of wire from coil L2.
Now, reconnect the B -plus lead to the plate
coil of the r.f. stage and tune in a weak signal
near the center of the desired tuning range.
Peak coils L3, L5, and L6. Coil L1 will tune
very broadly. Recheck the neutralization once
again (after removing the r.f. B -plus lead)
and secure the turns of coil L2 with a spot of
cellulose cement or colorless nail polish. As a
final check, measure the plate current of the
r.f. stage. It should run approximately 8 ma.
and should not vary when the antenna is disconnected from the stage. A variation in plate
current indicates oscillation of the r.f.
amplifier.
If a noise generator is available, coil L1 and
the antenna tap can be adjusted for a onedecibel or so improvement in noise figure
after the above adjustments are completed.
Adjustment of the six -meter converter is
identical to the above outline.
Plate power requirements of
each converter are 70 volts
at 8 ma. for the r.f. stage,
and 105 volts at approximately 10 ma. for
the mixer and oscillator. A suitable supply is
The Converter
Power Supply
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A
-I-
+
o
#
+
I
-Fink
Ì
+
-Iv
+
+v
I
H-
+ _IN
+
+
v
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Receivers and Transceivers
550
Figure 24
SCHEMATIC, "SIAMESE" CONVERTER FOR 2- AND 6- METERS
-5
-512
turns hookup wire close wound about
Id. trimmer capacitor.
the body of 6
L9 -19/s turns :32 e. close wound about the
trimmer
capacitor.
6
,o.fd.
of
body
L,0-16 turns :28 e., tap 51/2 turns from
ground. Some as 1.1.
L,1- Neutralizing coil. 50 turns :36 e., wound
turns :26 e., on 14 -inch
Coil data: LI, L3
diameter polystyrene rod or form. Wind
3/8 -inch long, top 2 turns from ground end
on L,. Adjust by spreading turns.
L2-Neutralizing coil. 20 turns :30 e. on
10 megohm resistor, close
5 :32" diameter
wound. Adjust by spreading turns.
L4
turn hookup wire over 8 -plus end of
coil L3.
L5-6 turns, some as L,. Tap 2 turns from
ground end.
L6, 116 -26 turns :32 e. on 1/4-inch slug -tuned
form, close wound. (Cambridge :PLS -6 v.h.f.
form with green colored slug.)
turns hookup wire over 8 -plus end of
L7, LIT
Coils L6 and L,6.
L8
-I
some as L2.
L,2 -19 turns :28 e., some construction os L6.
turn hookup wire over B -plus end of
L,3
coil L12.
L14 -Some as L,3, wound over ground end of
coil L15.
L,5 -17 turns :28, same construction as L6.
L18-25 turns :32 e., same construction as L6.
L19
turns hookup wire over 8 -plus end of
coil L16. See text.
-1
-2
-2
vl
6Cwa
vz
6AK5
A. F
L2
C2,500
MIA
L3
LI
L6
LS
O
4
/44-146 MC
°411.
Ca,
L
1= C3
MC.
C6
C 20
000
500
1000
500
2B-30
9
V3
6AK5
Yac
X1
38.66
MC
c
C7
1000
P,
Br 70V.
%
t
I
V4
6Cw4
Ln
O
Q
p
=
C23
500
GND.
I
6.3V.
2
Sz
500
MIA.
Lis
Lis
5
Cu
L13
Lu
°
50 -52 MC.
RCS.
51
50 MCI
6AK5
Li2
Lio
MC.I
Vs
A.F.
C12,500
8+105V
`
--o A-
IMF
-T-
-3
500
2 8-30 MC.
Re
2206
Cis
Cis
000
1000
NOTES
Ve
I. ALL RESISTORS i /2-WATT.
6AK5
2.500 ULF CAPACITORS ERIE
OSC.
GP
-500 SILVCR MICA BUTTON.
I1
3. 1000 LUF CAPACITORS, ERIE GP -500 SILVER MICA BUTTON.
4. 61.UF TRIMMER CAPACITORS CENrRALAB B2P -6 OR EOUIV.
CINCH-JONES l33-65 -10 -001.
5. NUVISTOR SOCKETS
6. FILAMENT BYPASS CAPACITORS: CENTRALAB DISC
L19
X2
22.0 MC.
0
HIR;
10501
C25
500
Cie
1000
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100K=
Re, 470
HANDBOOK
2 -6 M. "Siamese" Converter
of the triode -connected 6AK5 operating with
grid injection. The link circuit from the r.f.
amplifier stage is tapped directly to the 144
Mc. mixer coil to obtain optimum coupling.
The Local Oscillator Stage. A 6AK5 tube is
used as the crystal controlled oscillator in the
2 -meter converter. A 38.66 Mc. overtone
crystal oscillates in a grid- screen circuit, with
the plate circuit tuned to the third harmonic
(116 Mc.). The oscillator is capacitively
coupled to the grid circuit of the mixer stage.
The six -meter converter makes use of a
6AK5 overtone oscillator using a 22 Mc.
crystal. The tube is connected as a triode, and
the oscillator is inductively coupled to the
cathode circuit of the mixer stage. This configuration is required to obtain sufficient injection voltage without permitting the 22 Mc.
frequency to appear in the broadly tuned plate
circuit of the mixer. While the pentode mixer
is undoubtedly noisier than the triode, the
overall noise figure of the converter is much
less than the atmospheric noise at 50 Mc. so
this configuration does not tend to degrade
the usable sensitivity of the converter.
549
All chassis holes are drilled, the major components mounted in place, and then the auxiliary shields are soldered to the chassis. Placement of parts may be seen in figures 26 -29. The
six tube sockets lie along the center line of
the chassis and all wiring is done in a point to -point fashion. The 500 -µpfd. ceramic grid plate blocking capacitors pass through small
holes drilled in the interstage partitions and
are supported between the top terminal of the
r.f. stage plate coil and one lead of neutralizing
coil L_
The neutralizing coil, in turn, is
attached to the top (grid) terminal of the r.f.
stage grid coil. Every effort should be made
to make all leads in the r.f. stages as short and
direct as possible.
Mixer stage wiring is straightforward. The
cathode injection coil of the 50 Mc. mixer
may be made of a length of small hook -up
wire run from pin #2 of the 6AK5 socket,
looping twice around oscillator coil L1R, then
back to the 6AK5 socket, to be soldered to the
grounded filament terminal of the socket.
Lii.
Testing the
Wiring should be checked and
Converters
Converter
Construction
As each converter is extremely
small in size, it is simple to
construct both of them upon a
single chassis. The two units are therefore
mounted on a small copper plate measuring
3" x 7" in area, having a 1/2-inch turned down
lip running along the edges. A drilling template for the chassis is shown in figure 25.
If the sheet copper is not available, a phenolic
"printed circuit board" covered with a thin
layer of copper may be used as a substitute.
the mixer and oscilator tubes
placed in their sockets. Power
is applied to the converter and the oscillator
stage adjusted for operation. A grid -dip oscillator or a nearby receiver will serve as a handy
indicator of oscillation. You can temporarily
unground the 220K grid resistor of each mixer
stage and insert a low range micro- ammeter
in the circuit, tuning the oscillator controls
for maximum mixer grid current. If a v.t.m.
is handy, it may be attached to the grid pin of
the mixer stage and the oscillator controls
adjusted for maximum negative grid voltage.
Voltage should measure between
and
-1
Figure 23
THE "SIAMESE" CONVERTER
PROVIDES SUPERIOR V.H.F.
PERFORMANCE ON TWO BANDS
volts.
-2
Next, connect a receiver capable of tuning
the 28 -30 Mc. range to the output of the con-
This dual converter has a noise figure better
than 3 decibels on 2- and 6- meters. Utilizing
crystal control for maximum frequency stability and the new Nuvistor triode, superior
performance is achieved at minimugp cost. The
converter is built upon a small copper chassis
mounted to an aluminum panel by means of
two end plates. Panel size is 31/4" x 81/2 ".
Plate voltage is applied to both converters,
and filament voltage is controlled by the
panel mounted toggle switches. Nuvistor tube
(right) is compared to conventional 6AK5 in
foreground.
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548
THE RADIO
Receivers and Transceivers
Figure 21
EASE OF
MOUNTING
IN YOUR
AUTOMOBILE IS
FEATURED IN
THESE UNITS
The transceiver and
supply have low
that they may
be placed in line beneath
power
profile
so
the dashboard of your
automobile. Tuning controls are easily accessible
to driver of car.
of these converters is better than 3.5 decibels,
which compares favorably with units employing the expensive 417A low noise triode, and
may only be surpassed by use of the costly
416B tube.
For simplicity and ease of operation, the
two miniature converters are built on one
panel- chassis combination approximately 8" x
31/2" in size. The units may be powered from
the communications receiver, or may be run
from a separate supply as desired.
The circuits of the two con Description
verters are similar except for
minor details ( figure 24). A
6CW4 is used as a grid driven, neutralized
r.f. stage, link coupled to a 6AK5 mixer stage.
A second 6AK5 serves as a crystal -controlled
local oscillator. The intermediate frequency
range is 28 to 30 Mcs. The choice of a high
i.f. eliminates image problems and permits use
of a simple slug -tuned coupling circuit between the converter and the companion
Circuit
The Mixer Stage. A pentode- connected 6AK5
serves as a grid biased mixer for the 50 Mc.
converter. Cathode injection from the crystal
controlled local oscillator is used to achieve
proper mixing voltage. Use of a triode mixer
stage is not recommended as the reduction in
conversion noise of the triode over the pentode
is minimal at 50 Mc. and there is tendency of
the triode to regenerate as the frequency of
the injection oscillator is quite close to the intermediate frequency. The 144 Mc. mixer stage
takes advantage of the lower mixer noise level
Figure 22
THE RCA "NUVISTOR" VHF TUBE
The miniature RCA Nuvistor triode provides
high gain, low noise performance in the v.h.f.
spectrum at low cost. Intended for TV use,
this small tube shows excellent results in the
2- and 6 -meter converter described in this
section.
receiver.
The R.F. Stage. The r.f. stage of each converter
consists of a single 6CW4 Nuvistor triode.
Inductive neutralization is used (L2 and L11)
incorporating a series blocking capacitor to
remove plate voltage from the circuit. This
simple configuration provides above 20 decibels of usable gain, which is more than sufficient to override mixer noise. A single stage
such as this is noticeably less susceptible to
cross -modulation from strong local signals than
is a double stage (6BQ7A, for example) , or
two cascaded high gain stages.
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HANDBOOK
2 -6 M. "Siamese" Converter
milliameter across the plate "test"
points. A 20 watt lamp bulb may be attached
to the antenna receptacle as a dummy load.
Power is now applied and the pi- network circuit is adjusted for maximum glow of the
lamp. A 0 -10 d.c. milliameter placed across
the grid "test" points may be used to adjust
the excitation level to the 2E26. Grid current
should run between 2 and 3 ma., and plate
current is approximately 50 ma.
For 21 Mc. operation, the "grid tuning"
capacitor is resonated to 21 Mc. and the pinetwork retuned to this band. Slight adjustment of L3 and L4 will permit the two bands
to be properly tuned by swinging the resonating capacitors from minimum to maximum
capacitance.
The last step is to switch to v.f.o. operation,
and adjust the slug of coil L1 for proper dial
calibration. The slug should be permanently
fixed in position with a drop of nail polish
to prevent mechanical instability during mobile
operation of the unit.
The "magic eye" tube can be used to indicate amplifier resonance, but an external plate
meter is recommended for v.f.o. operation,
since loading must be readjusted as the transa 0 -100 d.c.
547
mitter frequency is varied. The "eye" tube
can be used for loading adjustment, but it
takes practice to interpret variations in the
pattern.
The Power Supply
An inexpensive power
supply suitable for a.c.
operation is shown in figure 20. Voltage regulation is employed for maximum stability.
Mobile supplies, such as the transistor types
shown in the Power Supply chapter are suitable for mobile operation. Low voltage required for operation of the v.f.o. and receiver
may be obtained from a dropping resistor and
regulator tube.
27 -5
"Siamese"
Converter for
Six and Two Meters
The new R.C.A. Nuvistor series of miniature tubes brings low noise level v.h.f. reception within the economic capability of the
average radio amateur. Described in this section are twin crystal controlled converters for
50 and 144 Mc. that make use of the 6CW4
Nuvistor v.h.f. triode. The inherent noise level
Figure 20
HOME MADE POWER
SUPPLY FOR
TRANSCEIVER FITS
IN MATCHING
CABINET WITH
SPEAKER
Simple transformer-oper ated a.c. supply is used
for home station work.
VR -ISO provides
regu-
lated voltage for maximum stability. Dynamic
speaker is included in
enclosure.
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546
THE RADIO
Receivers and Transceivers
speaker to the audio jack. Light the filaments
and apply plate voltage. Transformers T1, T2,
and T3 can be aligned by loosely coupling a
2050 kc. signal from an external source to the
plate circuit (pin #6) of the 6CG8 mixer
tube. Next, the bandswitch is placed in the 10
meter position and a 28 Mc. signal is applied
to the input circuit of the receiver. Proper
tracking is achieved in the usual manner, with
the oscillator padding capacitor determining
the calibration at the high frequency end of
the dial, and the variable slug of the oscillator
coil (L6) being used to set the edge of the
band at the low frequency end of the dial. An
Figure 19
REAR VIEW OF UNDER -CHASSIS
AREA
Point to point wiring is used, with many small
components soldered directly to the tube
socket pins. Coaxial antenna receptacle, microphone receptacle, and crystal- v.f.o. switch are
mounted on back apron of chassis. Pilot lamp
receptacles are bolted to frame of tuning
capacitor which is dropped below the chassis
deck by means of cut -out in deck.
antenna can now be connected to the antenna
jack of the transceiver and signals should be
heard. Mixer plate coil L, is peaked for maximum signal response near the center of the
band and adjustment of the transmitter pinetwork circuit can be made for greatest receiver sensitivity.
Bandswitch S2 is now placed in the 15
meter position and the oscillator padding capacitor is adjusted to correctly position the
high frequency end of the 15 meter band when
the tuning capacitor is at minimum setting.
The mixer padding capacitor is adjusted for
maximum receiver sensitivity at the same
frequency.
The transmitter portion should now be
aligned. Place the 6AU6 and 6CL6 tubes in
their respective sockets and insert a 7 Mc.
crystal in socket X2. Throw S1 to the transmit
position and adjust Ln for proper crystal oscillation. Next, plate coil L3 of the 6CL6 stage
is adjusted to 28 Mc. with the "grid tuning"
capacitor nearly open. Plate voltage is removed
and the 2E26 is inserted in its socket. Place
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HANDBOOK
Meter Transceiver
10 -15
of the stator support bars leaving them attached to one bar. Cut the rear plate so that
it is supported only by the other bar. A small
planetary unit is placed between the capacitor
and the dial for ease of tuning. A second planetary unit is used for the transmitter v.f.o.
All sockets, terminal strips, and trimmer
capacitors are mounted in place using 4 -40
hardware with soldering lugs placed .,eneath
the nuts in various convenient positions.
Transceiver
Wiring
The wiring of the unit is quite
simple if done in the proper
sequence. The under -chassis
area contains many small components but these
need not be crowded, provided proper care is
taken in the layout and installation of parts.
The smaller components (capacitors and resistors) are installed between the socket pins of
the various tubes. Socket ground connections
are made before the wiring is done, filament
wiring is done next, then the socket -mounted
components are placed in position. Number 22
stranded thermoplastic insulated wire (0.07"
diameter, Consolidated #737) is recommended
for all leads except the filament circuit. Number 18 wire should be used for these leads.
Small diameter, insulated phono- type" shielded wire is used for the lead running from
pin #2 of the 12AX7 to the receiver volume
control capacitor. The filament circuit is wired
in a series -parallel arrangement so that either
6- or 12 -volt operation may be chosen at the
power plug.
Before i.f. transformer T1 is mounted in
position, it should be modified so that it tunes
to 2050 kc. Some makes of transformers will
reach that frequency with no modification.
Others will require that some turns be removed
from the primary and secondary windings, or
that the value of internal fixed capacitance
be reduced accordingly.
The three small 15 meter variable padding
capacitors are mounted below the chassis in
close proximity to the bandswitch and may be
seen in the center of the chassis (figure 17)
Crystal socket X1 is mounted horizontally on
a small metal bracket under the rear of the
chassis so that the type FT -243 crystal may be
inserted and removed from the rear of the
transceiver. The buffer tuning capacitor
(marked "grid tuning" on the front panel) is
mounted on a small aluminum bracket at the
middle of the chassis. Oscillator coil L1 is posi-
'
.
-
545
REAR PANEL
REAR EDGE Or CHASSIS
Ta
xl
TUNING
T1
T -R
CAPACITOR
I
RDC6
SWITCH
6E5-M
I
PLATE
!TUNING/
1
DIRONT
ii
5UBPANEL
L FRONT
U LIRONT
PANEL
EDGE Or CABINET
Figure 18
LAYOUT OF MAJOR COMPONENTS
ABOVE THE CHASSIS
tioned between the buffer capacitor and the
v.f.o. capacitor, and is mounted in a small
aluminum shield cut down from an i.f. transformer can.
A dust plate is bolted to the rear lip of the
chassis and adds extra strength to the assembly by virtue of the two angle brackets bolted
to the plate and chassis. The 2E26 socket is
mounted on this plate, as are the v.f.o., crystal
switch, power plug, and antenna receptacle.
A slot is cut along the top edge of the dust
plate to insure adequate ventilation.
Transceiver Coils
Only six coils are required
for the transceiver, four of
them in the transmitter section. Because of the
compact construction and the influence of
nearby objects, it is wise to grid -dip each coil
to the proper frequency after installation.
Oscillator plate coil (L2) is resonated by the
internal capacitance of the associated tubes and
stray circuit capacitance, and should be grid dipped with the oscillator and buffer tubes in
their respective sockets.
Testing the
Transceiver
The transceiver should be tested
a section at a time. Start with
the receiver and audio portion.
Insert the tubes in the sockets and place crystal
X1 in the holder and connect a temporary
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544
THE RADIO
Receivers and Transceivers
tank circuit is very short. Directly behind the
tuning capacitor is the modulation transformer.
The 2E26 transmitting amplifier tube is
mounted in a horizontal position at the right
of the chassis, as shown in figure 15. The
transmitter pi- network output circuit is panel
mounted, directly in front of the plate cap of
the 2E26. The 6ME -10 tuning "eye" is panel
Figure 17
UNDERCHASSIS VIEW OF
TRANSCEIVER
V.f.o. tuning capacitor and coil (in shield) are
at top edge of chassis. "Grid" tuning capacitor
is recessed behind panel and driven with shaft
extension. The 21 Mc. padding capacitors are
directly behind bandswitch at center of chassis.
1.1. amplifier is along chassis edge in foreground.
mounted between the pi- network components
and the receiver tuning capacitor. Placement
of receiver components is conventional, with
the 6CG8 mixer stage mounted near the tuning
capacitor.
The receiver tuning capacitor is a two section unit, converted from a single section
Johnson 167 -3 variable capacitor. Using a
small hacksaw or jeweler's saw, the two stator
rods are cut so that a front group of plates are
supported by one post, and a rear plate is held
by the other post. This is the way you do this
operation: Leave the front two stator plates
and the rear stator plate in position, removing
the three other plates in between. Next, cut
the remaining two front plates away from one
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3
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542
Receivers and Transceivers
section. During the transmission the "eye"
indicates proper amplifier adjustment.
The three stage audio amplifier serves as a
modulator for the transmitter as well as an
audio system for the receiver. During transmission, both sections of a 12AX7 serve as a
voltage amplifier, driving two 6CM6 pentode
tubes in a parallel class A modulator circuit.
A simple resistive feedback circuit from the
plates of the modulator to the plate of the
driver stage improves speech quality and reduces distortion. The audio output transformer
T4 serves as a modulation choke when switch
section S1.1) opens the return circuit of the
loud speaker jack. Switch section SI_c couples
the modulator to the plate circuit of the r.f.
amplifier stage.
In the receiving mode, the audio signal
from the diode second detector circuit is applied through the volume control to the grid
circuit of the second section of the 12AX7
speech amplifier. The cathode circuit of the
first section of the 12AX7 is opened by switch
section S1.1) during reception.
Transceiver Layout
Figures 13, 15, 17 and
19 illustrate the general
plan of the transceiver.
The panel layout of controls is shown in figure
13, and parts placement above the chassis is
illustrated in figures 15 and 19. The transceiver
is built upon an aluminum chassis 67/8" x
55/8" x 1" in size. This assembly fits within a
steel wrap- around type cabinet .33,4" high,
63/8" deep and 7" wide. This cabinet was
custom -made to allow absolute minimum size
of the transceiver. A manufactured cabinet can
be used at a sacrifice in compactness. The California Chassis Co. type LTC-464 cabinet and
chassis, with an over -all measurement of 41/2"
x 91/8" x 71/8" is suitable and less expensive
than a custom package.
and Assembly
The transceiver makes use of a dual front
panel. Both panels are made of 1/16 inch
clear plastic sheet. The sub -panel is bolted
directly to the chassis and is painted black to
provide a good background for the tuning
dial. The front panel is a similar piece of
plastic, spaced about 1/4 inch in front of the
sub -panel by means of four bolts and metal
spacers. This panel is painted and lettered as
shown. For decorative purposes, a thin strip
of aluminum is run across the bottom of the
panel to provide a pleasing color contrast to
the eye.
Layout of principal parts above the chassis
can be observed by comparing the photographs
with figure 18. Viewed from the top front,
the receiver occupies the left portion of the
chassis and the transmitter occupies the right
half. The external plugs and receptacles are
mounted on the rear apron of the chassis.
The receiver tuning capacitor is centered on
the chassis, with the 6DC6 r.f. amplifier tube
mounted horizontally above it on a bracket.
The socket is oriented so that the grid connection between the tube and the amplifier
Figure 16
SCHEMATIC OF TRANSCEIVER
Receiver tuning capacitor-Oscillator section
3 -18 ppId. Detector section 3 -8 µµId. (See
text for details.)
Pi- network loading capacitor -400 µµId. Allied
Radio Co., Chicago, 111. #61 -H -009.
L,
1H. 1/2" diam. form, 1" long, tuned
with adjustable 1/4-20 iron core slug. (7.07.42 Mc.) Wind with # 18 e.
L2 -25 µh. Tunes to 7 Mc. with circuit capacitance. 5/16" diam., 1/2" long with adjustable 1/4 -20 iron core slug. Wind with
-1
22 e.
L3-0.9
pH. Tunes to 21 Mc. with tuning
capacitor at maximum, and 29.7 Mc. with
capacitor at minimum. 5/16" diam., 1/="
long with adjustable 1/4 -20 iron core slug.
Wind with »22 e.
L4-0.9 µh. Tunes to 21 Mc. with tuning
capacitor at maximum, and 29.7 Mc. with
capacitor at minimum. B&W coil, 3/4" diam.,
8 turns per inch #18 wire.
L5-1.5 µH. Tunes to 21 Mc. with tuning
capacitor at maximum and auxiliary padding capacitor in circuit, and 29.7 Mc. with
tuning capacitor at minimum and auxiliary
capacitor out of circuit.
-Two
windings. Tuned winding: 0.4 pH,
wound on 5/16" diam. form., 1/2" long with
adjustable 1/4 -20 iron core slug. Wind with
.-.18 e. Secondary winding: 0.4 pH scramble
wound, spaced 1/8 -inch from tuned winding,
#22 d.c.c. Tunes 25.95 -27.65 Mc. for 10
meters, 22.05 -22.5 Mc. for 15 meters.
Note: Coils may be wound on J. W. Miller Co.
#41 -A000 -CBI ceramic forms, with type R
slug. Alternatively, J. W. Miller Co. #20A
and #21A series adjustable r.f. coils may
be substituted. All coils should be adjusted
to frequency with o grid -dip oscillator.
T1 -2050 kc. i.?. transformer. J. W. Miller Co.
#13-WI. Remove turns from 1500 kc. windings to resonate at 2050 kc.
T2T3 -265 kc. i.f. transformer. J .W. Miller
L6
Co.
#12-H1.
74- Primary,
10
-watt.
5000 ohms. Secondary 4 ohms.
Dial -Made up of Jackson Bros. planetary
drive. (Arrow Electronics Co., 6S Cortland
St., New York 7, N.Y.)
Tuning Eye: 6ME -10 (midget) or EM -84. (See
tube manual for pin connections.)
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HANDBOOK
10 -15
semi - remote cutoff pentode is used as an r.f.
amplifier with the input grid connected directly to the r.f. circuit of the transmitter power
amplifier. Thus, when the transmitter is properly adjusted and loaded to the antenna system,
the receiver input circuit is automatically tuned
to the same frequency. This eliminates the
components and space normally required for a
tuned r.f. input circuit. The coupling capacitor and grid resistor of the 6DC6 stage are
chosen so that the tube blocks itself off during
transmission periods. The relatively large potential developed on the grid of the r.f. stage
does no harm. A 27 -ohm composition resistor
is placed in the plate lead of the r.f. amplifier
to suppress a parasitic oscillation that often
shows up in such circuitry. The resistor has no
effect upon the operation of the amplifier
stage.
Figure 15
OBLIQUE VIEW OF TRANSCEIVER
CHASSIS
The 2(26 power amplifier tube is mounted in
a horizontal position, supported by bracket at
rear of the chassis. Chassis is perforated below tube to permit passage of air around tube.
Pi- network components are in front of tube
cap. Antenna coaxial receptacle is at rear of
chassis on bracket. Construction of multiple
front panel may be seen in this view.
Meter Transceiver
541
A triode -pentode (6CG8) is used as the
first mixer stage. The triode section operates
as a "hot plate" oscillator 2050 kc. below the
signal frequency. Grid injection is used to the
pentode mixer section. Parallel padding capacitors are switched across the mixer and oscillator circuits in order to tune the 15 meter
band. A 6BE6 pentagrid tube is employed as
a second mixer from 2050 kc. to 265 kc. The
#1 grid acts as the anode of a cathode feedback crystal oscillator, with the degree of feedback controlled by a capacity bridge placed between #1 grid, cathode and ground.
A single 6BÁ6 provides sufficient i.f. gain
at 265 k.c., and two transformers produce excellent "skirt" selectivity and adjacent signal
separation. The r.f. stage, the second mixer,
and the i.f. stage are all controlled by the
a.v.c. circuit, operating from one -half of a
6AL5 tube. The second diode section of the
6AL5 acts as the second detector and automatic noise limiter. The a.n.l. circuit has a
very low distortion level, and is in the circuit
at all times. A 6ME -10 miniature "magic eye"
tube serves as a signal strength indicator,
operating from the a.v.c. line of the receiver
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THE RADIO
Receivers and Transceivers
540
V
--O
A-P AMP.
MULTIPLIER
VIAL 05C.
()MC)
(21ORjCMC)
MIC
VOL.
B.6300
ei 1'O V.
V.
Figure 14
BLOCK DIAGRAM OF TRANSCEIVER
External power supply is used, so transceiver may be operated from a.c. supply or from mobile
power pack. Tuning adjustments are accomplished by means of 6E5 miniature "magic eye." The
"eye" tube shown is the imported type 6ME -10 (Concord Electronics Co., 809 No. Cahuenga Blvd.,
Los Angeles 38, Calif.). The FM -type EM -84 may be substituted.
has arisen for a compact transceiver that will
work well either in the car or at the home
station. The unit described in this section has
been designed to meet this need.
This compact transceiver package covers the
10 and 15 meter bands, and employs a stable
superheterodyne receiver and a 20 watt a.m.
transmitter. The transmitter may be either
crystal controlled, or driven by the internal
v.f.o. A 10 watt audio system operates from
a crystal microphone and provides 100% modulation of the transmitter. During reception the
audio stages deliver sufficient power to drive
an external speaker well above the noise level
of the automobile.
Small enough to fit comfortably under the
dash of today's car, the transceiver delivers a
well modulated signal at a maximum plate
power load of 300 volts at 170 milliamperes
and 150 volts at 20 milliamperes.
Transceiver
Circuit
A block diagram of the trans ceiver circuit is shown in
figure 14. Twelve tubes are
employed, three in the transmitter section,
three in the audio portion, and six in the receiver section. Change-over from receive to
transmit is accomplished by a four -pole, two position switch (S1), mounted in the upper
left corner of the front panel (figure 13).
One section of this switch (Six) removes the
plate and screen voltage from the transmitter
amplifier stage, another section (S).p) transfers the low voltage from the transmitter exciter stages to the receiver circuits, a third
switch section (Sl.D) activates either the
speech amplifier or the loud speaker jack, and
the fourth section (S)_B) sensitizes the "magic
eye" indicator tube for reception.
The receiver portion employs a double conversion circuit to achieve maximum image
rejection with adequate selectivity. A 6DC6
www.americanradiohistory.com
HANDBOOK
10 -15
fairly broad as is the r.f. stage tuning although
the antenna trimmer will have to be repeaked
when going from one end of the 80 meter
band to the other. However, if the mixer
trimmer capacity has to be changed at the low
end of a band to obtain an increase in S -meter
reading, it means that the coil will have to be
altered. If the trimmer has to be increased in
capacity, it indicates that more inductance is
needed and the tap will have to be moved
farther up on the coil. Conversely, if the
trimmer has to be decreased in capacity, less
inductance is needed and the tap is moved
down on the coil. The Erie trimmer capacitors
pass from maximum to minimum capacity in
180 degrees of rotation and are at minimum
capacity when the lettering on the cap is
adjacent to the mounting bolts.
The final dial calibration must be made
with a bottom shield on the chassis
temporary piece of screen wire is satisfactory
in
order that the calibration will be correct when
the receiver is placed in the cabinet. 100 kc.
points are marked off on the dial from harmonics of the crystal calibrator and in between
points may be marked off with dividers since
the dial is fairly linear. The dial scale is removed for inking the calibration points and
when permanently reinstalled, it is covered
with a 1/16 inch piece of clear polystyrene
sheet the same size as the scale. This will keep
the dial clean and prevent warping of the
paper scale. A permanent pointer is made from
a long scrap of polystyrene sheet with an inked
-a -
Meter Transceiver
539
line scribed down the center of the pointer.
It can be shaped to fit over the planetary drive
by holding the plastic under hot water until it
is soft enough to be bent to shape.
The front panel is fastened to the chassis
by means of the hexagonal nuts holding the
bandswitch and toggle switches. Another 1/16
inch sheet of polystyrene or plexiglass is cut
to fit over the dial opening with the cutout
for the shaft of the planetary drive allowing
clearance for the tuning knob. Lettering decals
are used to mark the controls.
Any well filtered power supply that delivers
about 250 volts at 80 ma. is suitable for the
receiver. For 6 volt operation terminal 4 on
the power plug is jumpered to terminal 2
(ground) and power is applied to terminal 3.
For 12 volt operation terminal 3 is left open
and 12 volts applied to terminal 4.
27 -4
A Compact
Transceiver for
10 and 15 Meters
Regardless of "conditions" and the sunspot
cycle, the 10 and 15 meter bands are exceedingly popular with a large group of amateurs.
Many stations on these bands employ low
power, and the amateur using a low power
transmitter, inexpensive receiver and modest
antenna suffers no great handicap.
In addition, the growth of mobile operation
on these bands has been rapid and the need
..................
...
.......................41
. ...
.... 09400001,160000.0000000.0001000
00041"0004108000041000000061,100410000
.................................
..0..00....41..0.......0.000......
4000060.0.41.0410.0000.000000.0000000
0000.041414100041414141000000000000
....41.41...........41........0.0.....
...........................
41.41.000041414141410.0000041..414141000000.0.
0041.0.41414141.4100000.....0.0.....0
0.4141414141....0...00...41410.00000.00.
41041414141414100000.141000000410000004104100
0...410.0414141.0.41.041.0.0.00000...41
...041.41........
4141.......41.4141.4141.....ci
..
..
...........41.......... ..
.......4141.............
04104.40004100000.0000000000000000.0
Figure 13
POCKET -SIZE
TRANSCEIVER!
This
miniature transceiver is designed for
top-performance on the
21 and 28 Mc. amateur
bands. Receiver section
utilizes double conversion
for suppression of images
and for maximum selectivity. Modulated amplifier stage of transmitter
employs a 2E26. Power
supply and speaker are
contained in auxiliary
cabinet shown sitting
atop transceiver.
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538
THE RADIO
Receivers and Transceivers
The next step in the assembly of the receiver is to
mount the coil partitions with the attached
switch sections. The shaft is connected to the
switch index (previously mounted on the front
of the chassis) with a 1/4 -inch shaft coupling
and the switch index is rotated so that the
switch contacts are in proper alignment. The
nut holding the index can then be tightened
down to hold it in place.
The r.f. coils are wound as shown in the
coil table (figure 11) and the two grid coils
(L1 and L3) can be wired in without further
attention since they tune separately with the
antenna trimmer. The third set of contacts on
the grid coil switch wafer (SW1.c) are used
to switch in the additional 680 ohm cathode
resistor used to limit the excessive gain developed on the 80 meter band. This prevents
overloading the triode mixer with a strong
signal. The mixer and oscillator coils are next
wound and installed and the r.f. section is
ready to be tuned up.
Coil Assembly
proceeding with the
alignment of the oscillator and
mixer coils, a dial scale is made
up from stiff paper or white cardboard and
fastened to the dial back plate. The semicircular scales are made with a compass using
black india ink. A temporary pointer is made
of light aluminum or plastic scrap and is
attached to the planetary drive dial plate by
means of the small screws holding the dial
plate. The section of the pointer extending
over the scale is cut in half along the center
line so that only half of the pointer remains
to be used as a guide line for marking off the
calibration points on the dial. A rough alignment of the oscillator and mixer coils can be
made with a grid dip meter using the tuning
range data given in the coil table. The frequency of the oscillator circuit is higher than
the frequency of the mixer circuit by 3000 kc.
(the frequency of the i.f.) except in the case
of the 15 meter band where the oscillator
frequency is lower than the mixer frequency
by 3000 kc. As a starting point, the tuning
capacitor is set at almost full mesh (near maximum capacity) and each oscillator coil is set
at its lowest frequency by adjusting its associated trimmer to maximum capacity. The
mixer coils can be set in the same fashion on
Receiver
Calibration
Before
the low frequency edge of each amateur band.
The low band edges can be found by having
the receiver turned on for operation and applying a signal to the antenna input from a signal
generator or your transmitter v.f.o. The antenna trimmer will have to be peaked for each
band.
Once the low band edges have been found,
the rest of the alignment and dial calibration
is easily done using the built in 100 kc. calibrator. A short piece of wire is clipped to the
plate of the crystal calibrator tube and brought
near the antenna coils to get a fairly strong
signal to work with. A crystal harmonic should
fall at the low end of each band at the point
on the dial found previously with the external
signal. With this as a starting point, the tuning
capacitor can be rotated over the dial range
and 100 kc. points counted off to check the
coverage of each band on the dial. The coil
and capacitor combination given in the coil
table is designed to spread each band over
almost the entire 180 degrees of the dial using
a 15 µµfd. tuning capacitor. If an entire band
cannot be covered, the turns on the particular
oscillator coil must be squeezed together
slightly to increase the inductance and the
alignment procedure repeated by resetting the
oscillator trimmer so that the dial pointer
will align with the original calibration point
at the low end of the band. If the bandspread
is insufficient, the coil turns must be spread
to decrease the inductance. In the case of the
80 meter band it may mean removing one or
two turns from the oscillator coil.
When the band coverage has
been set by the above adjustment of the oscillator coils and
associated trimmers, the mixer coils are adjusted for proper tracking. Using the same
harmonics from the 100 kc. calibrator set the
tuning dial at the high frequency end of the
band, peak the antenna trimmer for a maxiTracking
Adjustments
mum reading on the S-meter and likewise peak
the trimmer capacitor on the mixer coil. Rotate
the tuning capacitor to the low frequency end
of the band and again adjust the same mixer
trimmer for a maximum reading on the Smeter. If no improvement can be made in the
meter reading, the mixer is tracking with the
oscillator and no further adjustment need be
made. The 80 and 40 meter mixer tuning is
www.americanradiohistory.com
HANDBOOK
Bandpass- Filter Receiver
537
Figure 12
iSWiTC
5.6n
SS
Mill
PLACEMENT OF
MAJOR
COMPONENTS
BENEATH THE
NOES'
120
UUF,
O
o
0
O0A2
I
5W3
6Ue
0
11
Le
6BZ6
o
Ol
C+
O
1.
I
e
6005
'lo
01:
SW
eue
0
BFO
PITCH
0
_OJ
The 6BZ6 r.f. tube socket
Tz
eTe
o
i°
LO_
located so that the
rear wall of the coil compartment passes over the
center of the socket, isolating the input and output circuits. The 6U8
mixer socket is mounted
in the some fashion with
the mixer pins (I, 8, and
9) falling within the coil
compartment and the
oscillator pins (2, 6, and
7) placed in the oscillator
coil area of the chassis.
Bandchange switch segments are mounted to
the walls of the coil compartment and are driven
by the switch index affixed to the front wall of
the chassis. The antenna
trimming capacitor and
b.f.o. pitch capacitor are
mounted to extension
"ears" on the rear wall
of the coil compartment.
is
I
:10
60.16
LO
CHASSIS
o
808
`iYi 5w2
L9
\--SHAFT COUPLING
_0J
of the ceramic trimmer capacitors and the coil
leads are connected to the same lugs, with a
heavy wire going to the respective terminals
on the band switch. There is a continuously
shorting deck on the oscillator switch section
that picks up the unused coils and shorts them
all together to prevent any absorption loss
from occurring in the higher frequency coils.
The two 3000 kc. i.f. transformers are made
from standard 1500 kc. units by removing 5
feet of wire from each winding of the transformer. The transformers shown are J. W.
Miller # 13 -W 1, but other makes of transformers could also be altered to tune to 3000 kc.
The proper frequency is easy to check with a
grid dip meter. The b.f.o. coil is made the easy
way by using a broadcast -type "vari- loopstick."
Twelve inches of wire is removed from the
coil and an additional 18 inches is unwound
to make the cathode tap and is then rewound
back on the coil. The end of the rewound
section of the coil will be the ground end. A
padding capacitor of 68 µµfd. is placed across
this coil and the modified " loopstick" is
mounted in a shield can to match the i.f.
transformers. The slug of the coil tunes the
circuit to the exact frequency and the variable
pitch control capacitor between cathode and
ground provides about 3 kc. variation each
side of zero beat.
Preliminary Checking
and Adjustment
Most of the wiring
can be done before
the coil partitions and
the r.f. coils are installed. At this point the
tubes can be put in the sockets (figure 12)
and power applied for a preliminary check on
the i.f. and audio circuits. Tuning the i.f.
channel is a simple matter because the center
frequency is determined by the bandpass filter.
A low level 3000 kc. signal from a signal
generator or grid dip meter is applied through
a capacitor to the input of the filter and the
slugs of the i.f. transformers are adjusted for
maximum signal reading on the S- meter. The
S -meter is adjusted with the "zero" control to
read zero with no signal and the meter sensitivity is determined by the value of the meter
ground resistor. The value of 47K used with
the 0 -1 ma. meter will give full scale deflection with a very strong local signal.
www.americanradiohistory.com
THE RADIO
Receivers and Transceivers
536
near that part of the circuit in which they are
used to allow short leads and to facilitate
wiring (figure 10) Terminal boards can be
made up from one inch wide strips of fiberglass sheet or phenolic material using soldering lugs and rivets for the terminals. A plain
piece of the same material is placed between
the terminal board and the side of the chassis
and the two pieces are fastened to the chassis
with 4 -40 hardware. The trimmer capacitors
for the mixer coils are mounted on the coil
partition so that their terminals extend into
the mixer coil section adjacent to the proper
coil. 4 -40 tapped holes in the lip of the partition allow the trimmers to be fastened directly to the partition. These trimmers can be
wired to the band switch before the partition
is mounted to the chassis, and the leads of
the coils may be soldered to the trimmer capacitor terminals when they are installed. Notice
that one terminal of the trimmer capacitors
returns to a ground strap allowing them to be
.
tuned with a metal screw driver without shorting the B -plus voltage appearing on the mixer
coils. The bottom leads of the mixer coils
return to a common bus wire run near the
chassis and terminating on an insulated tie point near the 80 meter coil.
Shielded wire (phonograph pickup type) is
used in the audio circuit to bring the leads
from the 6T8 socket to the volume control,
but all r.f. wiring is unshielded, short and
direct. B -plus leads, S -meter wiring and the
a.v.c. and r.f. gain control wires pass around
the inside edge of the chassis where they are
out of the way. The disc -type bypass capacitors
are mounted right at the tube sockets with
short leads. The metal center posts of the r.f.
and i.f. tube sockets are grounded and the
cathode bypass capacitors are positioned to
cross the center of the sockets to further isolate
the input and output circuits of the tubes.
The silver mica padding capacitors in the
oscillator circuit mount directly on the lugs
Figure 11
COIL DATA
All r.f. coils are wound
on 138" lengths of !'2" diameter polystyrene rod in a space of 2/4" except
coil Li which is close wound in 9 16" space. The antenna coils are wound at the ground end of
the grid coils and spaced 1 16" below the grid coil. All wire is plain enamel in the sizes shown.
Holes are drilled through the forms to fasten the end of the coils and the forms are tapped for
4 40 bolts at the bottom ends to attach them to the chassis. The tcp point is the number of
turns from the bottom (B -plus) end of the coil.
BAND
COIL
TUNING RANGE
80
L1
Grid 3500 -7300
Ant.
Mixer 3500 -4000
Osc. 6500 -7000
45
#30
17
30
30
26
Mixer 7000 -7300
Osc. 10 -10.3 Mc.
35
L11
L,;
Grid 14 -30 Mc.
L
L
Ant.
Mixer
13
10
L.,
L1_19
40
20
15
10
L,
TURNS
20
14
Lti
Mixer
Osc. 18 -18.5 Mc.
7 1/2
L,
Mixer 28 -30 Mc.
9
L1d
Osc. 31 -33 Mc.
B.f.o. coil
BC
51/2
-
6
7
L1s
-21.5 Mc.
6
16
14 -14.4 Mc.
Osc. 17 -17.4 Mc.
L1
WIRE
80
37
L12
21
TAP
"vari -loop tick"
6
7
(see
text)
www.americanradiohistory.com
TRIMMER FIXED PAD
µµtd.
ppfd.
C-1
8 -50
4 -30
26
20
8 -50
20
30
20
20
C-1
20
20
8 -50
20
20
8 -50
4 -30
8-50
4-30
4 -30
4 -30
None
85
120
110
30
HANDBOOK
Bandpass -Filter Receiver
SW6
0
535
POWER
PLUG
Figure 10
o
vTCRMI
TERMINAL BOARD
UNDER -CHASSIS
REAR APRON
LAYOUT AND
PLACEMENT OF
TERMINAL BOARDS
Terminal boards are
NAL.
BOARD
TERMINAL
BOARD
I
I
COIL
COMPARTMENT
I
L
I
J
LEFT SIDE
RIGHT SIDE
the dial which is 53/4" long and /8" high. It
has 3/8" lips and takes the form of a shallow
pan. Two holes are drilled in the bottom lip
of the pan to fasten it to the chassis flush with
the front edge. The center of the pan is in
line with the center of the chassis and the
capacitor shaft. Tapped 6 -32 holes in the
chassis make it easy to mount the dial pan.
The center hole for the planetary drive is
found by sliding the back plate of the dial up
against the capacitor shaft and marking the
location of the hole. The drive mechanism is
then positioned on the back plate and bolted
to it. After these parts are permanently
mounted, a couple of braces are affixed from
the back plate of the dial to the chassis so
that the whole dial assembly and tuning
capacitor are held rigid (figure 8).
The Coil Assembly
Light sheet aluminum
is used to make the
under -chassis partitions that separate the coils
and act as mounting plates for the bandswitch
sections. The rear partition holding the antenna switch section, the antenna trimmer and
the b.f.o. capacitor measures 71/2" x 17/8"
with 1/4 -inch lips bent at right angles to the
top and bottom (figure 9) . The lips provide
stiffening and permit easy mounting to the
chassis. The front partition holding the oscillator and mixer switch sections measures 51/2"
x 1 %s ", with the same 1/4 -inch lips top and
bottom. The mixer coil trimmers are mounted
on the lip and do not project beyond the depth
mounted on the left,
right, and rear side walls
of the chassis. A piece of
phenolic material placed
beneath the board insulates the terminals from
the chassis. R.f. bypass
capacitors are mounted
directly to the pins of
the tube sockets.
of the chassis. Two side pieces attach to the
partitions to hold them rigid. Threaded holes
are cut in the bottom lips of the partitions so
that they can be fastened to the chassis with
4 -40 self- tapping screws. Cutouts are made in
each partition to clear the r.f. and oscillator
tube sockets and two 1/4-inch feed -through
holes are drilled in one side partition for the
B -plus and plate leads to the r.f. stage.
The bandswitch wafers, SW1, SW2, and
SW3 are mounted with bolts and spacers on
the coil partitions on a center line directly
below the tuning capacitor. The antenna
trimmer capacitor and the b.f.o. pitch capacitor are mounted on the rear partition three
inches to each side of the band switch center
line. Fiber extension shafts bring the controls
up to the front panel. A flat sided fiber shaft
is passed through the band switch sections and
is connected to the switch index with a metal
shaft coupling. The switch index is mounted
on the front apron of the chassis. The location of the holes to be drilled in the panel is
found by placing the drilled chassis against
the panel and marking the holes from the inside of the chassis. The large cutout for the
dial and the S-meter can be made with a fine
toothed coping saw.
Wiring the
Three small phenolic terminal
boards are used to mount most
of the miscellaneous resistors and the audio coupling capacitors. The
boards are bolted to the side of the chassis
Receiver
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534
THE RADIO
Receivers and Transceivers
chassis by 6 -32 bolts in the space between
each capacitor. The ground terminals of the
coils, trimmers, and padding capacitors attach
to ground lugs affixed to the chassis at the
ground bolt of the trimmers. The tuning capacitor is mounted on a small L- shaped bracket
and is inverted to place the terminals and
ground strap closer to the chassis. The ground
strap on the capacitor projects through a hole
in the chassis and makes connection to the
ground terminal underneath the chassis. The
tuning capacitor is positioned along the center
line of the chassis at a distance from the front
panel determined by the length of the dial
drive mechanism.
For smooth tuning the planetary
drive must be lined up carefully
with the shaft of the tuning capacitor. The
drive mechanism mounts on the back plate of
The Dial
Figure 9
UNDER -CHASSIS VIEW OF RECEIVER
The general layout of components beneath the chassis is shown in this view. The bandswitch passes
through the central coil comportment. Each switch segment is mounted to a wall of the compartment. The two antenna coil forms are mounted behind the coil compartment with the primary
windings connected to the antenna receptacle by a short length of coaxial line. To the right is the
r.f. stage tuning capacitor, and to the left is the b.f.o. pitch capacitor.
The bank of mixer coils is centered in the compartment and the various trimming capacitors are
mounted on the top front flange of the compartment to facilitate adjustment. Note that the
rear wall of the compartment passes across the socket of the r.f. stage, thus isolating the input
and output circuits of the tube.
The bank of oscillator coils is mounted between the compertment and the front wall of the chassis,
with the oscillator switch mounted to the outside wall of the compartment. All wiring is short and
direct, and most small bypass capacitors are mounted directly to the tube socket pins. Resistors
are mounted on terminal boards placed on the walls of the chassis.
!
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Bandpass -Filter Receiver
up for in the following i.f. stage using standard transformers. The bandpass filter requires no special coupling circuits and is
symmetrical as viewed from its terminals
in other words, either terminal may serve as
input or output at a nominal impedance of
4700 ohms. Mixer plate voltage may be applied directly to the filter since it is tested
at a voltage far in excess of any value normally
encountered in receivers. The gain control is
in the cathode of the i.f. stage as well as the
r.f. stage for more effective control of the
over -all gain of the receiver. The second 6BJ6
i.f. tube has its screen voltage regulated for
more effective operation of the bridge -type
S -meter in its plate circuit.
-
The Detector and
The detector and audio
stages are conventional
and the noise limiter is
a series diode (part of the 6T8 tube) with a
fixed threshold level that does a good job of
limiting noise without causing apparent distortion on phone signals. A switch is included
to disable the limiter when it is not needed.
The audio volume control is isolated from d.c.
potentials by coupling capacitors to eliminate
the tendency of these controls to become noisy
Audio Stages
when used in current carrying circuits. A
standby switch opens the cathode of the 6AQ5
stage silencing the receiver, but other methods
can be used such as cutting the B -minus of
the separate power supply or relay switching
the B -plus of a mobile power supply.
Receiver
The 81/2" x 11" x 2" chassis
size is just right for building
this receiver and there is
plenty of room for all the components without
crowding, even if the exact parts specified are
not used. Location of the major components
is shown in the photographs and the layout
follows the circuit diagram (figure 7) , passing around the chassis with the r.f. section
taking up most of the center area. It is a good
idea to make paper templates for the band pass filter and the i.f. transformers, marking
the drilling holes on the chassis from the templates. The various tube sockets should be
oriented so that grid and plate leads do not
have to cross over the sockets. Oscillator
trimmer capacitors are mounted on top of the
chassis in a line above the oscillator coils with
the capacitor leads projecting through 1/4 -inch
holes to the under side of the chassis. The
oscillator coils can then be mounted under the
Construction
Figure 8
REAR VIEW OF
BAN DPASS- FILTER
RECEIVER
The audio stages, 100
kc. calibrator crystal, and
b.f.o. are located along
the left edge of the
chassis. Intermediate am-
plifier stages
533
pass across
the back of the chassis,
with the 3 Mc. bandpass
crystal filter and voltage
regulator tube at right.
Oscillator alignment capacitors are placed directly behind the homemade dial, with the main
tuning capacitor centered
on the chassis. The r.f.
and mixer tubes are at
the center of the chassis.
Along the rear apron of
the chassis are (I. to r.):
audio jacks and a.n.l.
switch, antenna receptacle, and power plug.
The 5 -meter "zero"-set
potentiometer is placed
atop the chassis, between
the 100 kc. crystal and
the first i.f. tube.
411
www.americanradiohistory.com
II
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www.americanradiohistory.com
E$
HANDBOOK
Bandpass -Filter Receiver
plug and antenna connector. The most -used
controls are on the front panel. The tuning
dial is a very smooth planetary type having a
long plastic pointer which extends over a calibrated scale. Indirect lighting of the dial is
provided by two panel bulbs recessed at the
edge of the dial scale. A direct reading S -meter
is connected in the plate circuit of the last
i.f. amplifier tube, and a 100 kc. calibrator
is included.
Receiver Circuit
The tube lineup consists of
a high gain 6BZ6 semi remote cutoff r.f. amplifier; a 6U8 triode
mixer and oscillator; two 6BJ6 i.f. amplifiers;
a 6T8 detector, noise limiter and audio amplifier; and a 6AQ5 audio output stage. A second
6U8 is used for a combined beat oscillator and
crystal calibrator, and an 0A2 serves as a
voltage regulator.
The circuit is conventional with several
simplifications that make for less work and
easier construction, to say nothing of fewer
parts (figure 7) . The main d.c. supply voltage to the plates and screens of the tubes is
divided by resistors to eliminate coupling
through a common voltage source so extra
decoupling networks are not required. A voltage regulator tube stabilizes the oscillator and
screen voltages as well as the voltage on the
b.f.o. Bandswitching for the five amateur bands
is accomplished with only three switch wafers
(SWIA,B,c, SW2A_B, and SW3) and the
main tuning dial is used only for the oscillator
circuit (C2B) and plate coil of the r.f. stage
(C.,A) . This calls for only a small inexpensive
two gang capacitor. Only two coils (L1 and
L3) are used in the r.f. grid circuit to cover
five bands. Coil L1 covers the 80 and 40
meter bands and coil L3 covers the 20, 15
and 10 meter bands, trimmed by the 100 µµEd.
antenna capacitor, C1. The r.f. tuning is fairly
broad and the trimmer only requires resetting
when going from one end of a band to the
other.
The plate circuit of the r.f. amplifier (C2A,
L5.0) is resonated using separate coils for
each band which are preadjusted to track with
the oscillator tuning. Except for the 80 meter
band, the plate coils are tapped for proper
oscillator tracking, eliminating the need for
extra series or padding capacitors required
with other tracking methods. The oscillator
531
circuit (C2B,L10 -L14) uses single winding
coils in a Colpitts arrangement utilizing large
padding capacitors which serve the dual purpose of stabilizing the oscillator as well as
providing proper bandspread for the tuning
ranges. The oscillator tuning capacitor C2B
is as small in capacity as can be used to cover
the desired tuning range. The oscillator circuit is designed as if it were going to be used
for a v.f.o. in a transmitter
which calls for
mechanical rigidity and use of short leads, a
ceramic switch and tube socket, silver mica
capacitors, and solidly mounted coils. The coils
are wound on polystyrene rods and are bolted
to the chassis. Directly above each coil (on
top of the chassis) are the ceramic trimmer
capacitors (C4 -Cg) used to adjust the tuning
range (figure 8) . The capacitor lugs project
through holes drilled in the chassis and fall
adjacent to their respective coils and switch
contacts. The trimmer capacitors have a negative temperature coefficient, and the combination of fixed silver mica padding capacitors
and the negative compensating characteristics
of the adjustable trimmers tends to stabilize
the oscillator frequency with respect to temperature changes. Ceramic capacitors, together
with the small, rigid plates of the tuning
capacitor make the oscillator almost immune
to vibration.
-
The Mixer Stage
A triode mixer is not
commonly used in band switching receivers but its low internal noise
and low plate resistance make it ideal for
working into the low impedance of the crystal
bandpass filter. The injection voltage from
the oscillator is fed directly into the cathode
of the triode section of the 6U8. Although the
injection voltage varies from one band to the
next, the value is not critical and is sufficient
on all bands. A v.t.v.m. reading at the grid
of the oscillator (pin 2) will show between
and
volts for proper operation. A
5600 ohm resistor is placed across the 80
meter oscillator coil (L10) to reduce the
injection voltage on this band to the proper
level.
-5
-8
The I.F. Amplifier
The crystal bandpass
filter (Blackhawk Engineering Co.) has a maximum insertion loss
of only 3 db. This loss is more than made
www.americanradiohistory.com
530
in the collector circuit. Optimum collector load
for the 2N217 is approximately 500 ohms,
and the 2N217 develops a maximum audio
signal of 75 milliwatts at this load impedance.
Transformer T, matches the transistor circuit
to the 12 ohm miniature loudspeaker. The
receiver draws a maximum signal current of
11 milliamperes from the 9 -volt battery supply. If no external antenna is used, the receiver should be moved about to orient the
"loopstick" coil L, for best pickup of each
individual broadcast station. Adjacent channel
interference can often be eliminated by careful rotation of the set to "null out" the offending signa:. Ample loudspeaker volume will be
obtained from local stations without the use
of an external antenna.
27 -3
THE RADIO
Receivers and Transceivers
An Inexpensive
Bandpass- Filter
Receiver
A very selective high performance amateur
band receiver can be built by using a high
frequency crystal bandpass- filter in the i.f.
system. Selectivity and image rejection are
accomplished without the complex circuitry
and elaborate construction required in a dual
conversion receiver to obtain the same results.
The intermediate frequency used in this receiver is 300 kc., yet the bandwidth is only
3 kc. at 6 db down and 12 kc. at 60 db down
with sharp skirt selectivity. The receiver covers
all amateur bands between 10 and 80 meters.
To supplement this efficient i.f. system, the
entire receiver has been designed towards the
goal of simplicity both in circuitry and mechanical construction without sacrificing anything in performance or leaving out any controls needed in a communication receiver. The
receiver is built on a standard size chassis with
a compact perforated cabinet suitable for use
in the shack, or in the car as a first rate mobile
receiver (figure 6) . The power supplies are
built as a separate unit for this purpose and
the tube filaments are wired in a series parallel configuration so that either a 6 or 12
volt d.c. supply may be used. The speaker is
external and the seldom -used noise limiter
switch and headphone jack are on the rear
apron of the chassis, along with the power
Figure 6
HIGH PERFORMANCE AMATEUR BAND RECEIVER
MAKES USE OF 3 MC. CRYSTAL I.F. FILTER
This simple, easy to build receiver achieves a near ultimate in selectivity and sensitivity
without the complex circuitry of a dual conversion receiver.
The bandspread dial is made up from an inexpensive vernier unit, to which a celluloid
pointer has been attached. The dial opening is covered with a thin sheet of lucite, held
in position with 4 -40 sheet metal screws. Directly below the tuning dial is the band switch with Vie antenna trimmer, the a.v.c. and b.f.o. switch, and the i.f. gain control
to the left. Above these is the S- meter, mounted to the rear of the panel. To the right
are the b.f.o. pitch control, the standby and calibration switches, and the audio
gain control.
The receiver sits on four rubber feet to prevent the operating table from being marred
by the metal case. The front panel is attached to the receiver chassis, and the ventilated
cabinet is bolted to the rear of the chassis. Receiver size is only 11" x 6 ".
www.americanradiohistory.com
HANDBOOK
529
Circuitry and
27 -1
Components
It is the practice of the editors of this
Handbook to place as much usable information
in the schematic illustration as possible. In
order to simplify the drawing the component
nomenclature of figure 1 is used in all the following construction chapters.
The electrical value of many small circuit
components such as resistors and capacitors is
often indicated by a series of colored bands
or spots placed on the body of the component.
Several color codes have been used in the past,
and are being used in modified form in the
present to indicate component values. The
most important of these color codes are illustrated in figure 2. Other radio components
such as power transformers, i -f transformers,
chokes, etc. have their leads color -coded for
easy identification as tabulated in figure 3.
27 -2
A Simple
Transistorized Portable
B -C
Receiver
Illustrated in figures 4 and 5 is an easy
to construct two transistor portable broadcast
receiver that is an excellent circuit for the
beginner to build. The receiver covers the
range of 500 kc. to 1500 kc. and needs no
external antenna when used close to a high
power broadcasting station. An external anten-
na may be added for more distant reception.
The receiver is powered from a single 9 -volt
miniature transistor battery and delivers good
speaker volume, yet draws a minimum of current permitting good battery life.
Circuit
Operation of the receiver may
be understood by referring to
the schematic diagram of figure
5. The tuned circuit L1 -C, resonates at the
frequency of the broadcasting station. A portion of the r -f energy is applied to the base of
the 2N112 p -n -p type transistor. A tapped
winding is placed on coil LI to achieve an impedance match to the low base impedance of
the transistor. Emitter bias is used on this
stage, and the amplified signal is capacitycoupled from the collector circuit to a 1N34
diode rectifier. The rectifier audio signal is
recovered across the 2K diode load resistor,
which takes the form of the audio volume
control of the receiver. The diode operates in
an untuned circuit, the selectivity of the receiver being determined by the tuned circuit
in the r -f amplifier stage.
The audio signal taken from the arm of the
volume control (R1) is applied to the base of
the 2N112 r-f amplifier which functions simultaneously as an audio amplifier stage. The
amplified audio signal is recovered across the
2K collector load resistor of the 2N112, and
is capacitively coupled to the base of a 2N217
p -n -p audio transistor. This stage is base and
emitter biased, having the output transformer
Description
Figure 5
INTERIOR VIEW
OF TRANSISTOR
RECEIVER
The speaker and output
transformer are mounted
at the left of the Masonite chassis. Top, center is the "loopstick" r-f
coil, and directly to the
right is the 10 millihenry
r -f choke in the collector
lead of the 2N112 transistor. Battery Is at lower
right.
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528
THE RADIO
Receivers and Transceivers
Transformer, choke and coil windings may be
damaged by incorrect wiring of the high -volt-
FIGURE 3
COMPONENT COLOR CODING
age leads.
POWER TRANSFORMERS
PRIMARY LEADS
The problem of meeting and overcoming
such obstacles is just part of the game. A true
radio amateur (as opposed to an amateur
broadcaster) should have adequate knowledge
of the art of communication. He should know
quite a bit about his equipment (even if purchased) and, if circumstances permit, he should
build a portion of his own equipment. Those
amateurs that do such construction work are
convinced that half of the enjoyment of the
hobby may be obtained from the satisfaction
of building and operating their own receiving
and transmitting equipment.
BLACK
IF rAPPEO
COMMON
BLACK
TAP
BLACK /YELLOW
END
BLACK
HIGH VOLTAGE WINDING
/ RED
RED
CENTER -TAP
RED /YELLOW
RECTIFIER FILAMENT WINDING- YELLOW
YELLOW /BLUE
CENTER -TAP
FILAMENT WINDING
N
GREEN
1
GREEN /YELLOW
CENTER -TAP
FILAMENT WINDING
N
BROWN
2
CENTER -TAP
FILAMENT WINDING
BROWN
N
/YELLOW
SLATE
3
SLATE /YELLOW
CENTER -TAP
The Transceiver
I-F TRANSFORMERS
PLATE LEAD
BLUE
0+ LEAD
RED
GRID (OR DIODE
LEAD
)
A -V -C (OR GROUND)
GREEN
LEAD
BLACK
AUDIO TRANSFORMERS
PLATE LEAD
(PR/.)
BLUE OR BROWN
(PR /. )
GRID LEAD (SEC.)
GRID RETURN (SEC.)
RED
BY LEAD
GREEN OR YELLOW
BLACK
circuits. If possible, the wiring should be
checked by a second party as a safety measure. Some tubes can be permanently damaged
by having the wrong voltages applied to their
electrodes. Electrolytic capacitors can be
ruined by hooking them up with the wrong
voltage polarity across the capacitor terminals.
A popular item of equipment on "five meters"
during the late "thirties," the transceiver is
making a comeback today complete with modern tubes and circuitry. In brief, the transceiver
is a packaged radio station combining the elements of the receiver and transmitter into a
single unit having a common power supply
and audio system. The present trend toward
compact equipment and the continued growth
of single sideband techniques combine naturally with the space-saving economies of the
transceiver. Various transceiver circuits for the
higher frequency amateur bands are shown in
this chapter. The experimenter can start from
these simple circuits and using modern miniature tubes and components can design and
build his complete station in a cabinet no larger than a pre -war receiver.
EXTERNAL
ANTENNA
10H
.002
B20
0.0.5
C
o.5
100 LFD
+
is
Rr
B
HIVOLUNE
A30
2 n
o.S
100LF0
+
15
BUS
Figure 4
SCHEMATIC OF TRANSISTOR BROADCAST RECEIVER
-volt transistor battery. RCA VS -300
-9
-365
Nµfd. Lafayette Radio Co. MS -214, or Allied Radio Co. 61H -009
"Loopstick" coil. Lafayette Radio Co. MS -166
loudspeaker, 12 -ohm voice coil. Lafayette Radio Co. SK-39
T,-500 ohm pri., 12 ohm sec. Transistor transformer. Thordarson TR -18
C,
L,- Transistor
LS
-3"
www.americanradiohistory.com
527
times such a comparison is surprising.
When the builder has finished the wiring of
a receiver it is suggested that he check his
wiring and connections carefully for possible
errors before any voltages are applied to the
experimenter's instinct, even in those individuals owning expensive commercial receivers.
These lucky persons have the advantage of
comparing their home -built product against the
best the commercial market has to offer. Some-
STANDARD COLOR CODE- RESISTORS AND CAPACITORS
AXIAL LEAD RESISTOR
BLACK
_
M' l
-.'G1
--
TOLERANCE
MULTIPLIER
TOLERANCE
i
00
MULTIPLIER
3
YELLOW
4
TOLERANCE
GREEN
S
5
.000
0.000
00.000
000.000
0,000.000
00,000.000
000.000,000
IS
e
VIOLET
7
7
GRAY
e
e
9
WHITE
9
COEFF
CURE
COEFFICIENT
IJIISIII
('' rT,
COEFF
L'JI)aL:L4aIl1
I- TOLERANCE
TOLERANCE
MULTIPLIER
LTC MULTIPLIER
MULTIPLIER
BY -PASS COUPLING CERAMIC CAPACITOR
I
1.11
OIIM1111I-
IGURE
LIST FIGURE
AXIAL LEAD CERAMIC CAPACITOR
CAPACITY
TEMP. COEFF.
CAPACITY
-MULTIPLIER
I
TEMPERATURE
5- ROTRADIAL LEAD CERAMICCAPACITOR EXTENDED RANGE TC CERAMIC HICAP
RADIAL LEAD (BAND) RESISTOR
Q1111
I
2
1ST FIGURE
TOLERANCE
CAPACITY
0
4
¡ MULTIPLIER
II
3-DOT
5 -DOT
NONE
2
3
RADIAL LEAD DOT RESISTOR
%1I
DOT COLOR
MULTIPLIER
1
BLUE
WIRE-WOUND RESISTORS NAVE 1ST
DIGIT BAND DOUBLE WIDTH.
DISC CERAMIC RMA CODE
THIRD RING
0
0
BROWN
RED
ORANGE
2ND SIGNIFICANT FIGS
I STa
SECOND RING
END COLOR
SECOND FIGURE
INSULATED FIRSTRING
BODY COLOR
UNINSULATED
FIRST FIGURE
COLOR
DROWN- INSULATED
BLACK - NON -INSULATED
V(OPT
`E
O I
NI!
TOLERANCE
MULTIPLIERS
TOLERANCE
MULTIPLIER
MOLDED MICA TYPE CAPACITORS
RMA 3 -DOT (OBSOLETE)
CURRENT STANDARD CODE
1
ST L SIGNIFICANT FIGURE
RATED 300 V.D.C.
±
BLACK(JAN)
®
CLASS
1srL
(
¡
2ND
'
JAN
o eljAULTIPLIER
MULTIPLIER
CODE
i____%7 SIGNIFICANT FIG.
"-TOLERANCE
-
RMA 6-DOT (OBSOLETE)
RMA 5 -DOT CODE (OBSOLETE)
sIa FIGURE WORK.
TOLERANCE
VOLT
MULTIPLIER
FRONT
In
t
/9OB RMA
EMULTIPLIER
WORK. VOLT.
REAR
WORK.
TOLERANCE
f
ND SIG. FIG.
1ST
I
,
CLASS
TOLERANCE¡F A 4s,L
2ND
WHITE (RMA)",....te
BUTTON SILVER MICA CAPACITOR
20 % TOL.
MULTIPLIER
1ST
,qp
l
MULTIPLIER
3RD DIGIT
RMA 4 -DOT (OBSOLETE)
} SIG. FIGURE
WORK. VOLTAGE
J
MULTIPLIER
MULTIPLIER
L L TOLERANCE
TOLERANCE
1ST DIGIT
2ND DIGIT
2NDJ
FIGURE
157 j SIG.
WORKING VOLTAGE
BLANK
VOLT.
MOLDED PAPER TYPE CAPACITORS
MOLDED FLAT CAPACITOR
TUBULAR CAPACITOR
NORMALLY STAMPED
FOR VALUE
COMMERCIAL CODE
r2NDjSIGNIFICANTFIGURE
MULTIPLIER
PrI LI
TOLERANCE
JAN CODE CAPACITOR
SILVER
1571
-MULTIPLIER
SIG. VOLTAGE FIG.
A 2 -DIGIT VOLTAGE RATING INDICATES
900 V. ADD 2 ZEROS TO END OF 2 DIGIT
1ST
l SIGNIFICANT
2ND1
FIG.
BOO
I
WI
1ST I
-WORKING VOLTS
BLACK
MORE THAN
2NDJ SIGNIFICANT
1ST J
FIGURE
MULTIPLIER
TOLERANCE
CHARACTERISTIC
NUMBER.
Figure 2
STANDARD COLOR CODE FOR RESISTORS AND CAPACITORS
The standard color code provides the necessary information required to properly identify color coded
resistors and capacitors. Refer to the color code for numerical values and the number of zeros (or multi-
plier) assigned to the colors used. A fourth color band on resistors determines the tolerance rating as
follows: Gold= 5%, silver -10%. Absence of the fourth band indicates a 20% tolerance rating.
Tolerance rating of capacitors is determined by the color code. For example: Red --2%, green =5%, etc.
The voltage rating of capacitors is obtained by multiplying the color value by 100. For example:
Orange =3 x100, or 300 volts.
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CHAPTER TWENTY -SEVEN
Receivers
Transceivers
and
Receiver construction has just about become
lost art. Excellent general coverage receivers
are available on the market in many price
ranges. However, even the most modest of
these receivers is relatively expensive, and most
of the receivers are designed as a compromise
-they must suit the majority of users, and
they must be designed with an eye to the price.
It is a tribute to the receiver manufacturers
that they have done as well as they have. Even
so, the c -w man must often pay for a high fidelity audio system and S -meter he never
uses, and the phone man must pay for the c -w
man's crystal filter. For one amateur, the receiver has too much bandspread; for the next,
too little. For economy's sake and for ease
a
of alignment, low -Q coils are often found in
the r -f circuits of commercial receivers, making the set a victim of cross -talk and overloading from strong local signals. Rarely does
the purchaser of a commercial receiver realize
that he could achieve the results he desires
in a home -built receiver if he left off the frills
and trivia which he does not need but which
he must pay for when he buys a commercial
product.
The ardent experimenter, however, needs
no such arguments. He builds his receiver
merely for the love of the game, and the thrill
of using a product of his own creation.
It is hoped that the receiving equipment to
be described in this chapter will awaken the
FIGURE
1
COMPONENT NOMENCLATURE
CAPACITORS:
1-
RESISTORS
1- RESISTANCE VALUES ARE STATED IN OHMS, THOUSANDS
VALUES BELOW 99011JFD ARE INDICATED IN UNITS.
EXAMPLE. 1SOJJmFD DESIGNATED AS IS0.
2 - VALUES ABOVE
9991J.UFD ARE INDICATED IN DECIMALS.
AS .00S.
EXAMPLE: OOSJJFD DESIGNATED
3-
4-
(K),
AND MEGOHMS (MI.
EXAMPLE,
270 OHMS = 270
4700 OHMS = 4.7 H
33,000 OHMS = 33 H
100.000 OHMS = 100 N OR 0.1
33.000,000 OHMS' 33 M
OF OHMS
OTHER CAPACITOR VALUES ARE AS STATED.
EXAMPLE: ,01JFD, 0.51J1JFD, ETC.
M
2- ALL
RESISTORS ARE 1 -WATT COMPOSITION TYPE UNLESS
OTHERWISE NOTED. WATTAGE NOTATION IS THEN INDICATED
BELOW RESISTANCE VALUE.
TYPE OF CAPACITOR IS INDICATED BENEATH THE VALUE
DESIGNATION.
SM = SILVER MICA
C = CERAMIC
EXAMPLE:
M' MICA
47 N
0.5
P' PAPER
EXAMPLE.
250
EXAMPLE'
6-
M
.001
INDUCTORS.
ELECTROLYTIC OR "FILTER
INDICATED BELOW CAPACITY DESIGNATION.
5- VOLTAGE RATING
CAPACITOR IS
.01
P
C
MICROHENRIES= JAI
OF
20
10
450
'
600.
MILLIHENRIES=
HENRIES=
25
10
THE CURVED LINE IN CAPACITOR SYMBOL REPRESENTS
THE OUTSIDE FOIL 'GROUND OF PAPER CAPACITORS.
THE NEGATIVE ELECTRODE OF ELECTROLYTIC CAPACITORS,
OR THE ROTOR OF VARIABLE CAPACITORS.
tII
II
MH
H
SCHEMATIC SYMBOLSI
OR
-
-LT
IL
CONDUCTORS JOINED
CONDUCTORS CROSSING
BUT NOT JOINED
526
www.americanradiohistory.com
CHASSIS GROUNO
Noise
HANDBOOK
Miscellaneous
There are several other potential noise sources on a passenger vehicle, but they do not necessarily
give trouble and therefore require attention
only in some cases.
The heat, oil pressure, and gas gauges can
cause a rasping or scraping noise. The gas
gauge is the most likely offender. It will cause
trouble only when the car is rocked or is in
motion. The gauge units and panel indicators
should both be by- passed with the 0.1 -µfd.
paper and 0.00l-pfd. mica or ceramic combination previously described.
At high car speeds under certain atmospheric
conditions corona static may be encountered
unless means are taken to prevent it. The receiving-type auto whips which employ a plastic ball tip are so provided in order to minimize
this type of noise, which is simply a discharge
of the frictional static built up on the car. A
whip which ends in a relatively sharp metal
point makes an ideal discharge point for the
static charge, and will cause corona trouble
at a much lower voltage than if the tip were
hooded with insulation. A piece of Vinylite
sleeving slipped over the top portion of the
whip and wrapped tightly with heavy thread
will prevent this type of static discharge under practically all conditions. An alternative
arrangement is to wrap the top portion of the
whip with Scotch brand electrical tape.
Generally speaking it is undesirable from
the standpoint of engine performance to use
both spark -plug suppressors and a distributor
suppressor. Unless the distributor rotor clearance is excessive, noise caused by sparking
of the distributor rotor will not be so bad but
what it can be handled satisfactorily by a
noise limiter. If not, it is preferable to shield
the hot lead between ignition coil and distributor rather than use a distributor suppressor.
Suppression
525
In many cases the control rods, speedometer
cable, etc., will pick up high- tension noise
under the hood and conduct it up under the
dash where it causes trouble. If so, all control rods and cables should be bonded to the
fire wall (bulkhead) where they pass through,
using a short piece of heavy flexible braid of
the type used for shielding.
In some cases it may be necessary to bond
the engine to the frame at each rubber engine
mount in a similar manner. If a rear mountéd
whip is employed the exhaust tail pipe also
should be bonded to the frame if supported by
rubber mounts.
Determining the source of certain types of noise is made
difficult when several things
are contributing to the noise, because elimination of one source often will make little or
no apparent difference in the total noise. The
following procedure will help to isolate and
identify various types of noise.
Ignition noise will be present only when the
ignition is on, even though the engine is turning over.
Generator noise will be present when the
motor is turning over, regardless of whether
the ignition switch is on. Slipping the drive
belt off will kill it.
Gauge noise usually will be present only
when the ignition switch is on or in the "left"
position provided on some cars.
Wheel static when present will persist when
the car clutch is disengaged and the ignition
switch turned off (or to the left position), with
the car coasting.
Body noise will be noticeably worse on a
bumpy road than on a smooth road, particularly at low speeds.
Locating
Noise Sources
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524
Mobile
THE
Equipment
of the measures may already have been taken
when the auto receiver was installed.
First either install a spark plug suppressor
on each plug, or else substitute Autolite resistor plugs. The latter are more effective than
suppressors, and on some cars ignition noise
is reduced to a satisfactory level simply by
installing them. However, they may not do an
adequate job alone after they have been in use
for a while, and it is a good idea to take the
following additional measures.
Check all high tension connections for gaps,
particularly the "pinch fit" terminal connectors widely used. Replace old high tension
wiring that may have become leaky.
Check to see if any of the high tension wiring is cabled with low tension wiring, or run
in the same conduit. If so, reroute the low tension wiring to provide as much separation as
practicable.
By-pass to ground the 6-volt wire from the
ignition coil to the ignition switch at each
end with a 0.1-pfd. molded case paper capacitor in parallel with a .001 -µfd. mica or ceramic, using the shortest possible leads.
Check to see that the hood makes a good
ground contact to the car body at several
points. Special grounding contactors are available for attachment to the hood lacings on cars
that otherwise would present a grounding
problem.
If the high -tension coil is mounted on the
dash, it may be necessary to shield the high
tension wire as far as the bulkhead, unless
it already is shielded with armored conduit.
static
is either static
by rotation
of the tires and brake drums, or is noise generated by poor contact between the front
wheels and the axles (due to the grease in the
bearings). The latter type of noise seldom is
caused by the rear wheels, but tire static may
of course be generated by all four tires.
Wheel static can be eliminated by insertion
of grounding springs under the front hub caps,
and by inserting "tire powder" in all inner
tubes. Both items are available at radio parts
stores and from most auto radio dealers.
Wheel Static
Wheel
electricity generated
Voltage Regulator
Hash
Certain voltage regulators
generate an objectionable
amount of
"hash" at
the
higher frequencies, particularly in the v -h-f
range. A large by-pass will affect the operation
of the regulator and possibly damage the
points. A small by -pass can be used, however,
without causing trouble. At frequencies above
the frequency at which the hash becomes objectionable (approximately 20 Mc. or so) a
small by -pass is quite effective. A 0.001 -µfd.
RADI
O
mica capacitor placed from the field terminal
of the regulator to ground with the shortest
possible leads often will produce sufficient
improvement. If not, a choke consisting of about 60 turns of no. 18 d.c.c. or bell wire
wound on a h -inch form can be added. This
should be placed right at the regulator terminal, and the 0.001-µfd. by -pass placed from
the generator side of the choke to ground.
"whine" often can
be satisfactorily suppressed
from 550 kc. to 148 Mc. simply by by- passing
the armature terminal to ground with a special
Generator Whine
Generator
"auto radio" by-pass of 0.25 or 0.5 pfd. in
parallel with a 0.001 -µfd. mica or ceramic capacitor. The former usually is placed on the
generator when an auto radio is installed, but
must be augmented by a mica or ceramic capacitor with short leads in order to be effective at the higher frequencies as well as on the
broadcast band.
When more drastic measures are required,
special filters can be obtained which are designed for the purpose. These are recommended for stubborn cases when a wide frequency
range is involved. For reception only over a
comparatively narrow band of frequencies.
such as the 10 -meter amateur band, a highly
effective filter can be improvised by connecting between the previously described parallel
by -pass capacitors and the generator armature terminal a resonant choke. This may consist of no. 10 enamelled wire wound on a suitable form and shunted with an adjustable trimmer capacitor to permit resonating the combination to the center of the frequency band
involved. For the 10 -meter band 11 turns close
wound on a one -inch form and shunted by a
3-30 µµfd. compression -type mica trimmer is
suitable. The trimmer should be adjusted experimentally at the center frequency.
When generator whine shows up after once
being satisfactorily suppressed, the condition
of the brushes and commutator should be
checked. Unless a by -pass capacitor has
opened up, excessive whine usually indicates
that the brushes or commutator are in need of
attention in order to prevent damage to the
generator.
Loose linkages or body or frame
joints anywhere in the car are
potential static producers when the car is in
motion, particularly over a rough road. Locating the source of such noise is difficult, and
the simplest procedure is to give the car a
thorough tightening up in the hope that the
offending poor contacts will be caught by the
procedure. The use of braided bonding straps
between the various sections of the body of
the car also may prove helpful.
Body Static
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HANDBOOK
Noise Suppression
523
When using a PE -103A, or any dynamotor
for that matter, it may be necessary to devote
660
PE-103A
one set of contacts on one of the control relays to breaking the plate or screen voltage
to the transmitter oscillator, If these are supplied by the dynamotor, because the output of
a dynamotor takes a moment to fall to zero
when the primary power is removed.
6 V. INPUT
300
60
400
100
150
200
250
300
OUTPUT CURRENT, MA.
Vehicular Noise
26-5
Suppression
Figure
11
APPROXIMATE OUTPUT VOLTAGE
LOAD CURRENT FOR A PE -103A
DYNAMOTOR
VS.
less the 12 -volt brushes will
last almost as long as the 6-volt brushes.
The reason that these particular dynamo tors
can be operated in this fashion is that there
At 150 ma. or
are two 6-volt windings on the armature, and
for 12 -volt operation the two are used in series
with both commutators working. The arrangement described above simply substitutes for
the regular 6-volt winding the winding and
commutator which ordinarily came into operation only on 12 -volt operation. Some operators
have reported that the regulation of the PE103A may be improved by operating both commutators in parallel with the 6-volt line.
The three wires now coming out of the dynamotor are identified as follows: The smaller
wire is the positive high voltage. The heavy
wire leaving the same grommet is positive 6
volts and negative high voltage. The single
heavy wire leaving the other grommet is negative 6 volts. Whether the car is positive or
negative ground, negative high voltage can be
taken as car -frame ground. With the negative
of the car battery grounded, the plate current
can return through the car battery and the armature winding. This simply puts the 6 volts
in series with the 500 volts and gives 6 extra
volts plate voltage.
The trunk of a car gets very warm in summer, and if the transmitter and dynamotor are
mounted in the trunk it is recommended that
the end housings be left off the dynamotor to
facilitate cooling. This is especially important in hot climates if the dynamotor is to be
loaded to more than 200 ma.
When replacing brushes on a PE -103A check
to see if the brushes are marked negative and
positive. If so, be sure to install them accordingly, because they are not of the same material. The dynamotor will be marked to show
which holder is negative.
Satisfactory reception on frequencies above
the broadcast band usually requires greater
attention to noise suppression measures. The
required measures vary with the particular vehicle and the frequency range involved.
Most of the various types of noise that may
be present in a vehicle may be broken down
into the following main categories:
(1) Ignition noise.
(2) Wheel static (tire static, brake static,
and intermittent ground via front wheel bearings).
(3)
tacts.
"Hash"
from
voltage regulator con-
(4) "Whine" from generator commutator segment make and break.
(5) Static from scraping connections between various parts of the car.
There is no need to suppress ignition noise
completely, because at the higher frequencies
ignition noise from passing vehicles makes
the use of a noise limiter mandatory anyway.
However, the limiter should not be given too
much work to do, because at high engine
speeds a noisy ignition system will tend to
mask weak signals, even though with the limiter working, ignition "pops" may appear to
be completely eliminated.
Another reason for good ignition suppression at the source is that strong ignition
pulses contain enough energy when integrated
to block the a-v -c circuit of the receiver, causing the gain to drop whenever the engine is
speeded up. Since the a-v -c circuits of the
receiver obtain no benefit from a noise dipper, it is important that ignition noise be suppressed enough at the source that the a-v -c
circuits will not be affected even when the
engine is running at high speed.
The following procedure
should be found adequate for
reducing the ignition noise of practically any
passenger car to a level which the dipper can
handle satisfactorily at any engine speed at
any frequency from 500 kc. to 148 Mc. Some
Ignition Noise
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522
Mobile
THE
Equipment
RING
TIP
11
PRESS -TO
SWITCH
SHELL
(GROUND)
fYI
MIRE
PLUG
OF
-TALK
FigUrr
IO
STANDARD CONNECTIONS FOR THE
PUSH -TO -TALK SWITCH ON A HAND.
HELD SINGLE- BUTTON CARBON
MICROPHONE
RADI
O
a time, and the average "on" time should not
be more than half the average "off" time.
The output voltage vs. current drain is
shown approximately in figure 11. The exact
voltage will depend somewhat upon the loss
resistance of the primary connecting cable
and whether or not the battery is on charge.
The primary current drain of the dynamotor
proper (excluding relays) is approximately 16
amperes at 100 ma., 21 amperes at 160 ma.,
26 amperes at 200 ma., and 31 amperes at 250
ma.
microphones on the surplus market use these
connections.
There is an increasing tendency among mobile operators toward the use of microphones
having better frequency and distortion characteristics than the standard single -button
type. The high- impedance dynamic type is
probably the most popular, with the ceramic crystal type next in popularity. The conventional crystal type is not suitable for mobile
use since the crystal unit will be destroyed
by the high temperatures which can be reached
in a closed car parked in the sun in the summer time.
The use of low -level microphones in mobile
service requires careful attention to the elimination of common -ground circuits in the microphone lead. The ground connection for the
shielded cable which runs from the transmitter
to the microphone should be made at only one
point, preferably directly adjacent to the grid
of the first tube in the speech amplifier. The
use of a low-level microphone usually will require the addition of two speech stages (a pentode and a triode), but these stages will take
only a milliampere or two of plate current, and
150 ma. per tube of heater current.
Because of its availability
on the surplus market at a
low price and its suitability
for use with about as powerful a mobile transmitter as can be employed in a passenger car
without resorting to auxiliary batteries or a
special generator, the PE -103A is probably
the most widely used dynamotor for amateur
work. Therefore some useful information will
be given on this unit.
The nominal rating of the unit is 500 volts
and 160 ma., but the output voltage will of
course vary with load and is slightly higher
with the generator charging. Actually the 160
ma. rating is conservative, and about 275 ma.
can be drawn intermittently without overheating, and without damage or excessive brush
or commutator wear. At this current the unit
should not be run for more than 10 minutes at
PE -103A Dynemotor Power Unit
Only a few of the components in the base
are absolutely necessary in an amateur mobile
installation, and some of them can just as well
be made an integral part of the transmitter if
desired. The base can be removed for salvage
components and hardware, or the dynamotor
may be purchased without base.
To remove the base proceed as follows:
Loosen the four thumb screws on the base
plate and remove the cover. Remove the four
screws holding the dynamotor to the base
plate. Trace the four wires coming out of the
dynamotor to their terminals and free the lugs.
Then these four wires can be pulled through
the two rubber grommets in the base plate when
the dynamotoris separated from the base plate.
It may be necessary to bend the eyelets in the
large lugs in order to force them through the
gromm et s.
Next remove the two end housings on the
dynamotor. Each is held with two screws. The
high -voltage commutator is easily identified
by its narrower segments and larger diameter.
Next to it is the 12 -volt commutator. The 6volt commutator is at the other end of the armature. The 12 -volt brushes should be removed
when only 6 -volt operation is planned, in order
to reduce the drag.
If the dynamotor portion of the PE-103A
power unit is a Pioneer type VS-25 or a Russell type 530- (most of them are), the wires to
the 12 -volt brush holder terminals can be cross
connected to the 6-volt brush holder terminals
with heavy jumper wires. One of the wires disconnected from the 12-volt brush terminals is
the primary 12-volt pigtail and will come free.
The other wire should be connected to the opposite terminal to form one of the jumpers.
With
this arrangement it is necessary only
to remove the 6-volt brushes and replace the
12 -volt brushes in case the 6-volt commutator
becomes excessively dirty or worn or starts
throwing solder. No difference in output voltage will be noted, but as the 12 -volt brushes
are not as heavy as the 6-volt brushes it is
not permissible vi draw more than about 150
ma. except for emergency use until the 6-volt
commutator can be turned down or repaired.
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HANDBOOK
Control
Do not attempt to control too many relays,
particularly heavy duty relays with large coils,
by means of an ordinary push -to-talk switch
on a microphone. These contacts are not designed for heavy work, and the inductive kick
will cause more sparking than the contacts on
the microphone switch are designed to handle.
It is better to actuate a single relay with the
push -to-talk switch and then control all other
relays, including the heavy duty contactor for
the dynamotor or vibrator pack, with this relay.
The procedure of operating only one relay
directly by the push -to-talk switch, with all
other relays being controlled by this control
relay, will eliminate the often -encountered difficulty where the shutting down of one item of
equipment will close relays in other items as
a result of the coils of relays being placed in
series with each other and with heater circuits.
A recommended general control circuit, where
one side of the main control relay is connected
to the hot 6 -volt circuit, but all other relays
have one side connected to ground, is illustrated in figure 9. An additional advantage
of such a circuit is that only one control wire
need be run to the coil of each additional relay, the other side of the relay coils being
grounded.
The heavy -duty 6 -volt solenoid -type contactor relays such as provided on the PE -103A
and used for automobile starter relays usually
draw from 1.5 to 2 amperes. While somewhat
more expensive, heavy -duty 6-volt relays of
conventional design, capable of breaking 30
amperes at 6 volts d.c., are available with
coils drawing less than 0.5 ampere.
When purchasing relays keep in mind that
the current rating of the contacts is not a fixed
value, but depends upon (1) the voltage, (2)
whether it is a.c. or d.c., and (3) whether the
circuit is purely resistive or is inductive. If
in doubt, refer to the manufacturer's recommendations. Also keep in mind that a dynamotor presents almost a dead short until the armature starts turning, and the starting relay
should be rated at considerably more than the
normal dynamotor current.
The most generally used microphone for mobile work is the
single -button carbon. With a
high -output-type microphone and a high-ratio
microphone transformer, it is possible when
"close talking" to drive even a pair of push pull 6L6's without resorting to a speech amplifier. However, there is a wide difference in
the output of the various type single button
microphones, and a wide difference in the amount of step up obtained with different type
microphone transformers. So at least one
Microphones
and
Circuits
speech stage usually is desirable.
One of the most satisfactory single button
PUSH- TO-TALK
SWITCH ON MIKE
r
PUSH -TO
RELAY
Circuits
521
-TALK
Ll
I
V
rY
ALTERNATE
CONTROL
SWITCH
MAIN POWER RECEIVER
MUTING
RELAY
RELAY
ANTENNA
CHANGEOVER
RELAY
ANY
OTHER
RELAYS
Figure 9
RELAY CONTROL CIRCUIT
Simplified schematic of the recommended
relay control circuit for mobile transmitters.
The relatively small push -to -talk relay is
controlled by the button on the microphone
or the communications switch. Then one of
the contacts on this relay controls the other
relays of t he transmitter; one side of the
coil of all the additional relays controlled
should be grounded.
microphones is the standard Western Electric
type F -1 unit (or Automatic Electric Co. equivalent). This microphone has very high output
when operated at 6 volts, and good fidelity
on speech. When used without a speech amplifier stage the microphone transformer should
have a 50 -ohm primary (rather than 200 or 500
ohms) and a secondary of at least 150,000
ohms and preferably 250,000 ohms.
The widely available surplus type T-17 microphone has higher resistance (200 to 500
ohmsl and lower output, and usually will require a stage of speech amplification except
when used with a very low power modulator
stage.
Unless an F -1 unit is used in a standard
housing, making contact to the button presents
somewhat of a problem. No serious damage will
result from soldering to the button if the connection is made to one edge and the soldering is done very rapidly with but a small amount of solder, so as to avoid heating the
whole button.
A sound-powered type microphone removed
from one of the chest sets available in the
surplus market will deliver almost as much
voltage to the grid of a modulator stage when
used with a high -ratio microphone transformer
as will an F -1 unit, and has the advantage of
not requiring button current or a "hash filter."
This is simply a dynamic microphone designed
for high output rather than maximum fidelity.
The standardized connections for a single button carbon microphone provided with push to -talk switch are shown in figure 10. Practically all hand-held military-type single -button
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THE
Mobile Equipment
520
RADIO
Construction and
26 -4
Installation of Mobile
Equipment
Figure
PI-
8
NETWORK ANTENNA COUPLER
The pi- network antenna coupler is particularly satisfactory for mobile work since the
coupler affords some degree of harmonic reduction, provides o coupling variation to
meet varying load conditions caused by frequency changes, and can cancel out reactance presented to the transmitter at the end
of the antenna transmission line.
For use of the coupler on the 3.9 -Mc. band
CI should hove a maximum capacitance of
about 250 µµfd., LI should be about 9 microhenrys (30 turns 1" dia. by 2" long), and
C2 may include a fixed and a variable element with maximum capacitance of about 1400
µµ/d. A 100 -µµfd. variable capacitor will be
suitable at C1 for the 14 -Mc. and 28 -Mc.
bonds, with o 350 -44fd. variable at C2. Inductor LI should have an inductance of about 2 microhenrys (11 turns 1" dia. by 1"
long) for the 14 -Mc. band, and about 0.8 mlcrohenry (6 turns 1" dia. by 1" long) for the
28 -Mc. band.
quency. Or, if the tapped type of coil is used,
taps are changed until the proper number of
turns for the desired operating frequency is
found. This procedure is repeated for the different bands of operation.
Feeding the
Center- Loaded
Antenna
After much experimenting it
has been found that the most
satisfactory method for feeding the coaxial line to the
base of a center- loaded antenna is with the
pi- network coupler. Figure 8 shows the basic
arrangement, with recommended circuit con-
stants. It will be noted that relatively large
values of capacitance are required for all bands
of operation, with values which seem particularly large for the 75-meter band. But reference to the discussion of pi- network tank circuits in Chapter 13 will show that the values suggested are normal for the values of impedance, impedance transformation, and operating Q which are encountered in a mobile installation of the usual type.
It is recommended that the following measures be taken when constructing mobile equipment, either transmitting or receiving, to ensure trouble -free operation over long periods:
Use only a stiff, heavy chassis unless the
chassis is quite small.
Use lock washers or lock nuts when mounting components by means of screws.
Use stranded hook -up wire except where r-f
considerations make it inadvisable (such as
for instance the plate tank circuit leads in a
amplifier). Lace and tie leads wherever
necessary to keep them from vibrating or flopping around.
Unless provided with gear drive, tuning capacitors in the large sizes will require a rotor
lock.
Filamentary (quick heating) tubes should
be mounted only in a vertical position.
The larger size carbon resistors and mica
capacitors should not be supported from tube
socket pins, particularly from miniature sockets. Use tie points and keep the resistor and
capacitor "pigtails" short.
Generally speaking, rubber shock mounts
are unnecessary or even undesirable with passenger car installations, or at least with full
size passenger cars. The springing is sufficiently "sott" that well constructed radio
equipment can be bolted directly to the vehicle
without damage from shock or vibration. Unless shock mounting is properly engineered
as to the stiffness and placement of the shock
mounts, mechanical- resonance "amplification"
effects may actually cause the equipment to
v -h -f
be
shaken more than
if
the equipment were
bolted directly to the vehicle.
Surplus military equipment provided with
shock or vibration mounts was intended for
use in aircraft, jeeps, tanks, gun- firing Naval
craft, small boats, and similar vehicles and
craft subject to severe shock and vibration.
Also, the shock mounting of such equipment
is very carefully engineered in order to avoid
harmful resonances.
To facilitate servicing of mobile equipment,
all interconnecting cables between units
should be provided with separable connectors
on at least one end.
The send -receive control circuits of a mobile installation
dictated by the design of the equipment,
therefore will he left to the ingenuity of
reader. However, a few generalizations
suggestions are in order.
Control Circuits
are
and
the
and
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HANDBOOK
Mobile
A more effective radiator and a better line
match may be obtained by making the whip
approximately 10% feet long and feeding it
with 75 -ohm coax (such as RG-11 /U) via a
series capacitor, as shown in figure 6. The
relay and series capacitor are mounted inside
the trunk, as close to the antenna feedthrou h
or base-mount insulator as possible. The 10foot length applies to the overall length from
the tip of the whip to the point where the lead
in passes through the car body. The leads inside the car (connecting the coaxial cable,
relay, series capacitor and antenna lead)
should be as short as possible. The outer conductor of both coaxial cables should be grounded to the car body at the relay end with short,
heavy conductors.
100 -µµtd. midget variable capacitor is
suitable for C1. The optimum setting should
A
be determined experimentally at the center of
the band. This setting then will be satisfactory over the whole band.
One suitable coupling arrangement for either
a 1/4-wave or 5/16 -wave whip on 10 meters is
to use a conventional tank circuit, inductively
coupled to a "variable link" coupling loop
which feeds the coaxial line. Alternatively, a
pi- network output circuit may be used. If the
input impedance of the line is very low and
the tank circuit has a low C/L ratio, it may
be necessary to resonate the coupling loop
with series capacitance in order to obtain sufficient coupling. This condition often is encountered with a %-wave whip when the line
length approximates an electrical half wave-
length.
If an all -band center -loaded mobile antenna
is used, the loading coil at the center of the
antenna may be shorted out for operation of
the antenna on the 10 -meter band. The usual
type of center-loaded mobile antenna will be
between 9 and 11 feet long, including the center- loading inductance which is shorted out.
Hence such an antenna may be shortened to
an electrical quarter-wave for the 10 -meter
band by using a series capacitor as just discussed. Alternatively, if a pi- network is used
in the plate circuit of the output stage of the
mobile transmitter, any reactance presented
at the antenna terminals of the transmitter by
the antenna may be tuned out with the pi -network.
The All -Bond
Center-Loaded
Mobile Antenna
The great majority of mobile
operation on the 14 -Mc. band
and below is with center
loaded whip antennas. These
antennas use an insulated bumper or body
mount, with provision for coaxial feed from
the base of the antenna to the transmitter, as
shown in figure 7.
The center -loaded whip antenna must be
Antennas
519
CAR BODY
UNSHIELDED
LOADING COIL
RG-56/U LINE
TO
TRANSMITTER
COAXIAL LINE
GROUNDED TO
FRAME OF CAR
ADJACENT
TO BASE
OF
Figure
ANTENNA
7
THE CENTER -LOADED
WHIP ANTENNA
The center -loaded whip antenna, when provided with a topped loading coil or a series
of coils, may be used over a wide frequency
range. The loading coil may be shorted for
use of the antenna on the 10 -meter bond.
tuned to obtain optimum operation on the desired frequency of operation. These antennas
will operate at maximum efficiency over a
range of perhaps 20 kc. on the 75 -meter band,
covering a somewhat wider range on the 40meter band, and covering the whole 20 -meter
phone band. The procedure for tuning the antennas is discussed in the instruction sheet
which is furnished with them, but basically
the procedure is as follows:
The antenna is installed, fully assembled,
with a coaxial lead of RG-58 /U from the base
of the antenna to the place where the transmitter is installed. The rear deck of the car
should be closed, and the car should be parked
in a location as clear as
possible of trees,
buildings, and overhead power lines. Objects
within 15 or 20 feet of the antenna can exert
a considerable detuning effect on the antenna
system due to its relatively high operating Q.
The end of the coaxial cable which will plug
into the transmitter is terminated in a link of
3 or 4 turns of wire. This link is then coupled
to a grid -dip meter and the resonant frequency
of the antenna determined by noting the frequency at which the grid current fluctuates.
The coils furnished with the antennas normally are too large for the usual operating frequency, since it is much easier to remove
turns than to add them. Turns then are removed,
one at a time, until the antenna resonates at
the desired frequency. If too many turns have
been removed, a length of wire may be spliced
on and soldered. Then, with a length of insulating tubing slipped over the soldered joint,
turns may be added to lower the resonant fre-
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518
Mobile Equipment
THE
950
RADIO
COAX TO
RI
7511 COAX TO XuTR
Figure 6
RADIATOR FOR 10
METERS
If a whip antenna is made slightly longer
than one-quarter wave it acts as a slightly
better radiator than the usual quarter -wave
whip, and it can provide a better match to
the antenna transmission line if the reactance is tuned out by a serles capacitor close
to the base of the antenna. Capacitor C1 may
be a 100 -µµid. midget variable.
5/16 -WAVE
WHIP
remarks are in order on the subject of feed
and coupling systems.
Figure
5
A CENTER LOADED 80 -METER WHIP
USING AIR WOUND COIL MAY BE USED
WITH HIGH POWERED TRANSMITTERS
An anti -corona loop is placed at the top
of the whip to reduce loss of power and
burning of tip of antenna. Number of turns
in coil is critical and adjustable, high -Q
coil is refommended. Whip may be used
over frequency range of about 15 kilo-
cycles without retuning.
26 -3
Antennas for
Mobile Work
The most popular mobile anretna for 10 -meter operation
is a rear -mounted whip approximately 8 feet long, fed with coaxial line. This
is a highly satisfactory antenna, but a few
10 -Meter
Antennas
Mobile
The feed point resistance of a resonant quarter -wave rear -mounted whip is approximately
20 to 25 ohms. While the standing -wave ratio
when using 50 -ohm coaxial line will not be
much greater than 2 to 1, it is nevertheless
desirable to make the line to the transmitter
exactly one quarter wavelength long electrically at the center of the band. This procedure
will minimize variations in loading over the
band. The physical length of RG -8 /U cable,
from antenna base to antenna coupling coil,
should be approximately 5 feet 3 inches. The
antenna changeover relay preferably should be
located either at the antenna end or the transmitter end of the line, but if it is more convenient physically the line may be broken anywhere for insertion of the relay.
If the same rear -mounted whip is used for
broadcast -band reception, attenuation of broadcast -band signals by the high shunt capacitance of the low impedance feed line can be
reduced by locating the changeover relay right
at the antenna lead in, and by running 95 -ohm
coax (instead of 50 or 75 ohm coax) from the
relay to the converter. Ordinarily this will produce negligible effect upon the operation of
the .converter, but usually will make a worthwhile improvement in the strength of broadcast band signals.
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One -Tube
HANDBOOK
auto -set combination has not proven very satisfactory. The primary reason for this is the fact
that the relatively sharp i -f channel of the auto
set imposes too severe a limitation on the stability of the high- frequency oscillator in the
converter. And if a crystal- controlled beating
oscillator is used in the converter, only a portion of the band may be covered by tuning the
auto set.
The most satisfactory arrangement has been
found to consist of a separately mounted i.f.,
audio, and power supply system, with the converter mounted near the steering column. The
i -f system should have a bandwidth of 30 to
100 kc. and may have a center frequency of
10.7 Mc. if standard i -f transformers are to be
used. The control head may include the 144 Mc. r -f, mixer, and oscillator sections, and
sometimes the first i-f stage. Alternatively,
the control head may include only the h -f oscillator, with a broadband r -f unit included
within the main receiver assembly along with
the i.f. and audio system. Commercially manufactured kits and complete units using this
general lineup are available.
An alternative arrangement is to build a
converter, 10.7 -Mc. i -f channel, and second
detector unit, and then to operate this unit in
conjunction with the auto -set power supply,
audio system, and speaker. Such a system
makes economical use of space and power
drain, and can be switched to provide normal
broadcast-band auto reception or reception
through a converter for the h -f amateur bands.
A recent development has been the VHF
transceiver, typified by the Gonset Communicator. Such a unit combines a crystal controlled transmitter and a tunable VHF receiver
together with a common audio system and
power supply. The complete VHF station may
be packaged in a single cabinet. Various
forms of VHF transceivers are shown in the
construction chapters of this Handbook.
26-2
tion.
A
Converter
517
total transmitter power drain of about
80 watts from the car battery (6 volts at 13
amperes, or 12 volts at 7 amperes) is about
the maximum that can be allowed under these
conditions. For maximum power efficiency it
is recommended that a vibrator type of supply
be used as opposed to a dynamotor supply,
since the conversion efficiency of the vibrator
unit is high compared to that of the dynamotor.
A second school of thought states that the
mobile transmitter should be of relatively high
power to overcome the poor efficiency of the
usual mobile whip antenna. In this case, the
mobile power should be drawn from a system
that is independent from the electrical system
of the automobile. A belt driven high voltage
generator is often coupled to the automobile
engine in this type of installation.
variation of this idea is to employ
complete secondary power system in the
car capable of providing 115 volts a.c. Shown
in figure 4 is a Leece -Neville three phase
alternator mounted atop the engine block, and
driven with a fan belt. The voltage regulator
and selenium rectifier for charging the car
battery from the a -c system replace the usual
d -c generator. These new items are mounted
in the front of the car radiator. The alternator
provides a balanced deltaovtput circuit wherein the line voltage is equal to the coil voltage, but the line current is N/3 times the coil
current. The coil voltage is a nominal 6- volts,
RMS and three 6.3 volt 25 ampere filament
transformers may be connected in delta on
the primary and secondary windings to step
the 6-volts up to three -phase 115-volts. If
A
a
desired, a special
115 -volt, 3 -phase
transformer may be wound which will
less space than the three filament
formers. Since the ripple frequency of
phase d-c power supply will be quite
single
10 mfd
step -up
occupy
transa three
high, a
filter capacitor will suffice.
Mobile Transmitters
As in the case of transmitters for fixed -station operation, there are many schools of
thought as to the type of transmitter which is
most suitable for mobile operation. One school
states that the mobile transmitter should have
very low power drain, so that no modification
of the electrical system of the automobile will
be required, and so that the equipment may be
operated without serious regard to discharging
the battery when the car is stopped, or overloading the generator when the car is in mo-
Figure 4
LEECE -NEVILLE 3 -PHASE
ALTERNATOR IS ENGINE DRIVEN
BY
AUXILIARY FAN BELT.
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516
Mobile
converter end or the set end of the cable between the converter and receiver. This auxiliary trimmer should have a range of about 3
to 50 µµfd., and may be of the inexpensive
compression mica type.
with the trimmer cut out and the converter
turned off (by- passed by the "in -out" switch),
peak the regular antenna trimmer on the auto
set at about 1400 kc. Then turn on the converter, with the receiver tuned to 1500 kc., switch
in the auxiliary trimmer, and peak this trimmer
for maximum background noise. The auxiliary
trimmer then can be left switched in at all
times except when receiving very weak broadcast band signals.
Some auto sets, particularly certain General
Motors custom receivers, employ a high -Q high impedance input circuit which is very critical
as to antenna capacitance. Unless the shunt
capacitance of the antenna (including cable)
approximates that of the antenna installation
for which the set was designed, the antenna
trimmer on the auto set cannot be made to hit
resonance with the converter cut out. This is
particularly true when a long antenna cable is
used to reach a whip mounted at the rear of the
car. Usually the condition can be corrected by
unsoldering the internal connection to the antenna terminal connector on the auto set and
inserting in series a 100 -µµfd. mica capacitor.
Alternatively an adjustable trimmer covering
at least 50 to 150 µµfd. may be substituted for
the 100 -µµfd. fixed capacitor. Then the adjustment of this trimmer and that of the regular antenna trimmer can be juggled back and forth
until a condition is achieved where the input
circuit of the auto set is resonant with the converter either in or out of the circuit. This will
provide maximum gain and image rejection
under all conditions of use.
Reducing Battery
Drain of the
Receiver
THE
Equipment
When the
receiving installa-
tion is used frequently, and
particularly when the receiver is used with the car
parked, it is desirable to keep the battery
drain of the receiver -converter installation at
an absolute minimum. A substantial reduction
in drain can be made in many receivers, without appreciably affecting their performance.
The saving of course depends upon the design of the particular receiver and upon how
much trouble and expense one is willing to go
to. Some receivers normally draw (without the
converter connected) as much as 10 amperes.
In many cases this can be cut to about 5 amperes by incorporating all practicable modifications. Each of the following modifications
is applicable to many auto receivers.
If the receiver uses a speaker with a field
coil, replace the speaker with an equivalent
PM type.
RADIO
Practically all 0.3- ampere r-f and a -f voltage amplifier tubes have 0.15-ampere equivalents. In many cases it is not even necessary
to change the socket wiring. However, when
substituting i -f tubes it is recommended that
the i -f trimmer adjustments be checked. Generally speaking it is not wise to attempt to
substitute for the converter tube or a -f power
output tube.
If the a-f output tube employs conventional
cathode bias, substitute a cathode resistor of
twice the value originally employed, or add
an identical resistor in series with the one
already in the set. This will reduce the B
drain of the receiver appreciably without seriously reducing the maximum undistorted output. Because the vibrator power supply is much
less than 100 per cent efficient, a saving of
one watt of B drain results in a saving of nearly 2 watts of battery drain. This also minimizes the overload on the B supply when the
converter is switched in, assuming that the
converter uses B voltage from the auto set.
If the receiver uses push -pull output and if
one is willing to-accept a slight reduction in
the maximum volume obtainable without distortion, changing over to a single ended stage
is simple if the receiver employs conventional
cathode bias. Just pull out one tube, double
the value of cathode bias resistance, and add
a 25 -{ád. by -pass capacitor across the cathode
resistor if not already by- passed. In some
cases it may be possible to remove a phase
inverter tube along with one of the a -f output
tubes.
If the receiver uses a motor driven station
selector with a control tube (d-c amplifier),
usually the tube can be removed without upsetting the operation of the receiver. One then
must of course use manual tuning.
While the changeover is somewhat expensive, the 0.6 ampere drawn by a 6X4 or 6X5
rectifier can be eliminated by substituting six
115-volt r -m -s 50 -ma. selenium rectifiers (such
as Federal type 402D3200). Three in series
are substituted for each half of the full -wave
rectifier tube. Be sure to observe the correct
polarity. The selenium rectifiers also make a
good substitution for an OZ4 or OZ4 -GT which
is causing hash difficulties when using the
converter.
Offsetting the total cost of nearly $4.00 is
the fact that these rectifiers probably will last
for the entire life of the auto set. Before purchasing the rectifiers, make sure that there is
room available for mounting them. While these
units are small, most of the newer auto sets
employ very compact construction.
For reception on the 144 amateur band, and
those higher in frequency, the simple converterTwo -Meter Reception
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Mc.
Mobile Receiver Installation
HANDBOOK
KEEP THESE LEADS SHORT
OR SHIELD THEM.
\
VIBRATOR
I
SPD T.
6v
DC
TO HOT SIDE OF
VOICE COIL WINDING
515
mitter oscillator is killed instantly, thus avoiding trouble from dynamotor "carry over."
The efficiency of this arrangement is good
because the current drain on the main high
voltage supply for the modulated amplifier and
modulator plate(s) is reduced by the amount
of current borrowed from the receiver. At least
80 ma. can be drawn from practically all auto
sets, at least for a short period, without damage.
VIA CONTROL
RELAY IN XMTR.
6 V
Figure
3
METHOD OF ELIMINATING THE BATTERY
DRAIN OF THE RECEIVER VIBRATOR
PACK DURING TRANSMISSION
If the receiver chassis has room for o midget s.p.d.t. relay, the above arrangement not
only silences the receiver on transmit but
saves several amperes battery drain.
If the normally open contact on the relay is
connected to the hot side of the voice coil
winding as shown in figure 3 (assuming one
side of the voice coil is grounded in accordance with usual practice), the receiver will
be killed instantly when switching from receive to transmit, in spite of the fact that the
power supply filter in the receiver takes a
moment to discharge. However, if a "slow
start" power supply (such as a dynamotor or
a vibrator pack with a large filter) is used with
the transmitter, shorting the voice coil probably will not be required.
An alternative and high ly recommended procedure is to make use of
the receiver B supply on
transmit, instead of disabling it. One disadvantage of the popular PE -103A dynamotor is
the fact that its 450 -500 volt output is too high
for the low power r -f and speech stages of the
transmitter. Dropping this voltage to a more
suitable value of approximately 250 volts by
means of dropping
is wasteful of
power, besides causing the plate voltage on
the oscillator and any buffer stages to vary
widely with tuning. By means of a midget 6Using the Receiver
Plate Supply
On Transmit
resistors
volt s.p.d.t. relay mounted in the receiver, connected as shown in figure 2, the B supply of
the auto set is used to power the oscillator
and other low power stages (and possibly
screen voltage on the modulator). On transmit
the B voltage is removed from the receiver
and converter, automatically silencing the receiver. When switching to receive the trans-
It will be noted that with the arrangement of
figure 2, plate voltage is supplied to the audio
output stage at all times. However, when the
screen voltage is removed, the plate current
drops practically to zero.
The 200 -ohm resistor in series with the normally open contact is to prevent excessive
sparking when the contacts close. If the relay
feeds directly into a filter choke or large capacitor there will be excessive sparking at
the contacts. Even with the arrangement shown,
there will be considerable sparking at the contacts; but relay contacts can stand such sparking quite a while, even on d.c., before becoming worn or pitted enough to require attention.
The 200 -ohm resistor also serves to increase
the effectiveness of the .01 -fifd. r -f by -pass
capacitor.
One other modification of
the auto receiver which may
or may not be desirable depending upon the circumstances is the addi-
Auxiliary Antenna
Trimmer
tion of an auxiliary antenna trimmer capacitor.
If the converter uses an untuned output circuit
and the antenna trimmer on the auto set is
peaked with the converter cut in, then it is
quite likely that the trimmer adjustment will
not be optimum for broadcast -band reception
when the converter is cut out. For reception
of strong broadcast band signals this usually
will not be serious, but where reception of
weak broadcast signals is desired the loss in
gain often cannot be tolerated, especially in
view of the fact that the additional length of
antenna cable required for the converter installation tends to reduce the strength of
broadcast band signals.
If the converter has considerable reserve
gain, it may be practicable to peak the antenna
trimmer on the auto set for broadcast -band reception rather than resonating it to the converter output circuit. But oftentimes this results in insufficient converter gain, excessive
image troubles from loud local amateur stations, or both.
The difficulty can be circumvented by incorporation of an auxiliary antenna trimmer
connected from the "hot" antenna lead on the
auto r e c ei v e r to ground, with a switch in
series for cutting it in or out. This capacitor
and switch can be connected across either the
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Mobile
514
Equipment
(
6V6
THE
OPTIONAL
Br r0
6AQ5
OR
CONVERTER
1
OVC
Br
TO REST
OF SET
6X4
REGULAR
R -C FILTER
=OLF
30
200
..UF
2w
B+ 200
TO 250 V.
TO xMTR
-÷7°'
I
- -- 1
0 HY
V. VIA CONTROL
RELAY IN XMTR
6
I
I
I
FROM
RECEIVER
.i T
((}}
Br
I
""((}}
Ti
O
Br
TO LOW POWERI
SPEECH
STAGES
1
_
L
I
I
XMTR
J
I
Figure 2
USING THE RECEIVER PLATE SUPPLY
FOR THE TRANSMITTER
This circuit silences the receiver on transmit, and in addition makes it possible to use
the receiver plate supply for feeding the ex.
citer and speech amplifier stages In the
transmitter.
is to mount a small receptacle on the receiver
cabinet or chassis, making connection via a
matching plug. An Amphenol type 77 -26 receptacle is compact enough to fit in a very small
space and allows four connections (including
ground for the shield braid). The matching plug
is a type 70 -26.
To avoid the possibility of vibrator hash
being fed into the converter via the heater and
plate voltage supply leads, it is important that
the heater and plate voltages be taken from
points well removed from the power supply portion of the auto receiver. If a single -ended
audio output stage is employed, a safe place
to obtain these voltages is at this tube socket,
the high voltage for the converter being taken
from the screen. In the case of a push -pull output stage, however, the screens sometimes are
fed from the input side of the power supply
filter. The ripple at this point, while sufficiently low for a push -pull audio output stage,
is not adequate for a converter without additional filtering. If the schematic shows that
RADIO
the screens of a push -pull stage are connected
to the input side instead of the output side of
the power supply filter (usually two electrolytics straddling a resistor in an R -C filter), then
follow the output of the filter over into the r -f
portion of the set and pick it up there at a convenient point, before it goes through any additional series dropping or isolating resistors,
as shown in figure 2.
The voltage at the output of the filter usually runs from 200 to 250 volts with typical converter drain and the motor not running. This
will increase perhaps 10 per cent when the
generator is charging. The converter drain will
drop the B voltage slightly at the output of the
filter, perhaps 15 to 25 volts, but this reduction is not enough to have a noticeable effect
upon the operation of the receiver. If the B
voltage is higher than desirable or necessary
for proper operation of the converter, a 2 -watt
carbon resistor of suitable resistance should
be inserted in series with the plate voltage
lead to the power receptacle. Usually something between 2200 and 4700 ohms will be
found about right.
Receiver Disabling
on Transmit
When the battery drain is
high on transmit, as is the
case when a PE -103A dynamotor is run at maximum rating and other
drains such as the transmitter heaters and auto
headlights must be considered, it is desirable
to disable the vibrator power supply in the receiver during transmissions. The vibrator
power supply usually draws several amperes,
and as the receiver must be disabled in some
manner anyhow during transmissions, opening
the 6 -volt supply to the vibrator serves both
purposes. It has the further advantage of introducing a slight delay in the receiver recovery,
due to the inertia of the power supply filter,
thus avoiding the possibility of a feedback
"yoop" when switching from transmit to re-
ceive.
To avoid troubles from vibrator hash, it is
best to open the ground lead from the vibrator
by means of a midget s.p.d.t. 6 -volt relay and
thus isolate the vibrator circuit from the external control and switching circuit wires. The
relay is hooked up as shown in figure 3: Standard 8- ampere contacts will be adequate for
this application.
The relay should be mounted as close to
the vibrator as practicable. Ground one of the
coil terminals and run a shielded wire from
the other coil terminal to one of the power receptacle connections, grounding the shield at
both ends. By -pass each end of this wire to
ground with .01 pfd., using the shortest possible leads. A lead is run from the corresponding terminal on the mating plug to the control
circuits, to be discussed later.
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HANDBOOK
Mobile
shown. !Multi- position tone controls tied in
with the second detector circuit often permit
excessive "leak through." Hence it is recommended that the tone control components be
completely removed unless they are confined
to the grid of the a -f output stage. If removed,
the highs can be attenuated any desired amount
by connecting a mica capacitor from plate to
screen on the output stage. Ordinarily from
.005 to .01 tfd. wilt r:ovide a good compromise between fidelity and reduction of background hiss on weak signals.
Usually the s wit c h SW will have to be
mounted some distance from the noise limiter
components. If the leads to the switch are over
approximately 1 ii in c h e s long, a piece of
shield braid should be slipped over them and
grounded. The same applies to the "hot" leads
to the volume control if not already shielded.
Closing the switch disables the limiter. This
may be desirable for reducing distortion on
broadcast reception or when checking the intensity of ignition noise to determine the effectiveness of suppression measures taken on
the car. The switch also permits one to check
the effectiveness of the noise clipper.
The 22,000 -ohm decoupling resistor at the
bottom end of the i -f transformer secondary is
not critical, and if some other value already
is incorporated inside the shield can it may be
left alone so long as it is not over 47,000
ohms, a common value. Higher values must be
replaced with a lower value even if it requires
a can opener, because anything over 47,000
ohms will result in excessive loss in gain.
There is some loss in a -f gain inherent in this
type of limiter anyhow (slightly over 6 db),
and it is important to minimize any additional
loss.
It is important that the total amount of capacitance in the RC decoupling (r -f) filter not
exceed about 100 µµfd. With a value much
greater than this "pulse stretching" will occur
and the effectiveness of the noise clipper will
be reduced. Excessive capacitance will reduce
the amplitude and increase the duration of the
ignition pulses before they reach the clipper.
The reduction in pulse amplitude accomplishes
no good since the pulses are fed to the clipper
anyhow, but the greater duration of the lengthened pulses increases the audibility and the
blanking interval associated with each pulse.
If a shielded wire to an external dipper is employed, the r-f by -pass on the "low" side of
the RC filter may be eliminated since the capacitance of a few feet of shielded wire will
accomplish the same result as the by -pass
capacitor.
The switch SW is connected in such a manner that there is practically no change in gain
with the limiter in or out. If the auto set does
not have any reserve gain and more gain is
Receivers
513
needed on weak broadcast signals, the switch
can be connected from the hot side of the volume control to the j unction of the 22,000,
270,000 and 1 megohm resistors instead of as
shown. This will provide approximately 6 db
more gain when the clipper is switched out.
Many late model receivers are provided with
an internal r -f gain control in the cathode of
the r -f and /or i -f stage. This control should
be advanced full on to provide better noise
limiter action and make up for the loss in audio
gain introduced by the noise clipper.
Installation of the noise clipper often detunes the secondary of the last i -f transformer.
This should be repeaked before the set is permanently replaced in the car unless the trimmer is accessible with the set mounted in
place.
Additional clipper circuits will be found in
the receiver chapter of this llandbook.
Selectivity
While not of serious concern on
10 meters, the lack of selectivity
exhibited by a typical auto receiver will result
in QRM difficulty on 20 and 75 meters. A typical auto set has only two i -f transformers of
relatively low -Q design, and the second one
is loaded by the diode detector. The skirt selectivity often is so poor that a strong local
will depress the a.v.c. when listening to a
weak station as much as 15 kc. different in
frequency.
One solution is to add an outboard i-f stage
employing two good quality double -tuned transformers (not the midget variety) connected
"back -to- back" through a small coupling capacitance. The amplifier tube (such as a 6BA6)
should be biased to the point where the gain
of the outboard unit is relatively small (1 or
2), assuming that the receiver already has adequate gain. If additional gain is needed, it may
be provided by the outboard unit. Low- capacitance shielded cable should be used to couple
into and out of the outboard unit, and the unit
itself should be thoroughly shielded.
Such an outboard unit will sharpen the nose
selectivity slightly and the skirt selectivity
greatly. Operation then will be comparable to
a home -station communications receiver,
though selectivity will not be as good as a
receiver employing a 50 -kc. or 85 -kc. "Q5'er."
Obtaining Power
for the Converter
While the set is on the
bench for installation of
the noise clipper, provision should be made for obtaining filament and
plate voltage for the converter, and for the exciter and speech amplifier of the transmitter,
if such an arrangement is to be used. To permit removal of either the converter or the auto
set from the car without removing the other, a
connector should be provided. The best method
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512
Mobile
Equipment
THE
sary, and it is recommended that a noise clipper be installed without confirming the necessity therefor. It has been found that quiet reception sometimes may be obtained on 75 meters simply by the use of resistor type plugs,
but after a few thousand miles these plugs
often become less effective and no longer do
a fully adequate job. Also, a noise clipper in-
RADIO
TO A F.
POWER
AMP.
I
F
TRANS
sures against ignition noise from passing
trucks and "un- suppressed" cars. On 10 meters a noise clipper is a "must" in any case.
There are certain things that
should be done to the auto set
when it is to be used with a
converter, and they might as well be done all
at the same time, because "dropping" an auto
receiver and getting into the chassis to work
on it takes quite a little time.
First, however, check the circuit of the auto
receiver to see whether it is one of the few
receivers which employ circuits which complicate connection of a noise clipper or a converter. If the receiver is yet to be purchased,
it is well to investigate these points ahead of
Modifying the
Auto Receiver
time.
If the receiver uses a negative B resistor
strip for bias (as evidenced by the cathode of
the audio output stage being grounded), then
the additional plate current drain of the converter will upset the bias voltages on the various stages and probably cause trouble. Because the converter is not on all the time, it
is not practical simply to alter the resistance
of the bias strip, and major modification of the
receiver probably will be required.
The best type of receiver for attachment of
a converter and noise clipper uses an r -f stage;
permeability tuning; single unit construction
(except possibly for the speaker); push button
tuning rather than a tuning motor; a high vacuum rectifier such as a 6X4 (rather than an OZ4
or a synchronous rectifier); a 6SQ7 (or miniature or Loctal equivalent) with grounded cathode as second detector, first audio, and a.v.c.;
power supply negative grounded directly (no
common bias strip); a PM speaker (to minimize
battery drain); and an internal r -f gain control
(indicating plenty of built -in reserve gain
which may be called upon if necessary). Many
current model auto radios have all of the foregoing features, and numerous models have most
of them, something to keep in mind if the set
is yet to be purchased.
noise limiter either may be
built into the set or purchased
as a commercially manufactured unit for "outboard" connection via shielded wires. If the
receiver employs a 6SQ7 (or Loctal or miniature equivalent) in a conventional circuit, it is
a simple matter to build in a noise clipper by
Noise Limiters
A
Figure
1
SERIES -GATE NOISE LIMITER FOR AUTO
RECEIVER
Auto receivers using a 6SQ7, 786, 7X7, or
6A T6 as second detector and a.v.c. can be
converted to the above circuit with but few
wiring changes. The circuit hos the advantage of not requiring an additional tube socket for the limiter diode.
substituting a 6S8 octal, 7X7 Loctal, or a 6T8
9 -pin miniature as shown in figure 1. When
substituting a 6T8 for a 6ÁT6 or similar 7 -pin
miniature, the socket must be changed to a
miniature type. This requires reaming
the socket hole slightly.
If the receiver employs cathode bias on the
6SQ7 (or equivalent), and perhaps delayed
a.v.c., the circuit usually can be changed to
the grounded -cathode circuit of figure 1 without encountering trouble.
Some receivers take the r -f excitation for
the a -v -c diode from the plate of the i -f stage.
In this case, leave the a.v.c. alone and ignore
the a -v-c buss connection shown in figure 1
(eliminating the 1- megohm decoupling resistor).
If the set uses a separate a -v -c diode which
receives r-f excitation via a small capacitor
connected to the detector diode, then simply
change the circuit to correspond to figure 1.
In case anyone might be considering the use
of a crystal diode as a noise limiter in conjunction with the tube already in the set, it
might be well to point out that crystal diodes
perform quite poorly in series -gate noise clippers of the type shown.
It will be observed that no tone control is
9 -pin
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CHAPTER TWENTY -SIX
Mobile Equipment
Design and Installation
Mobile operation is permitted on all amateur
bands. Tremendous impetus to this phase of
the hobby was given by the suitable design of
compact mobile equipment. Complete mobile
installations may be purchased as packaged
units, or the whole mobile station may be home
built, according to the whim of the operator.
The problems involved in achieving a satisfactory two -way installation vary somewhat
with the band, but many of the problems are
common to all bands. For instance, ignition
noise is more troublesome on 10 meters than
on 75 meters, but on the other hand an efficient antenna system is much more easily accomplished on 10 meters than on 75 meters.
Also, obtaining a worthwhile amount of trans-
mitter output without excessive battery drain
is a problem on all bands.
26 -1
Mobile Reception
When a broadcast receiver is in the car, the
most practical receiving arrangement involves
a converter feeding into the auto set. The advantages of good selectivity with good image
rejection obtainable from a double conversion
superheterodyne are achieved in most cases
without excessive "birdie" troubles, a com-
difficulty with a double conversion superheterodyne constructed as an integral receiver
in one cabinet. However, it is important that
the b-c receiver employ an r -f stage in order to
provide adequate isolation between the converter and the high frequency oscillator in the
b -c receiver. The r -f stage also is desirable
from the standpoint of image rejection if the
converter does not employ a tuned output circuit (tuned to the frequency of the auto set,
usually about 1500 kc.). A few of the late
model auto receivers, even in the better makes,
do not employ an r -f stage.
The usual procedure is to obtain converter
plate voltage from the auto receiver. Experience has shown that if the converter does not
draw more than about 15 or at most 20 ma. total plate current no damage to the auto set or
loss in performance will occur other than a
slight reduction in vibrator life. The converter
drain can be minimized by avoiding a voltage
regulator tube on the converter h -f oscillator.
On 10 meters and lower frequencies it is possible to design an oscillator with sufficient
stability so that no voltage regulator is required in the converter.
With some cars satisfactory 75-meter operation can be obtained without a noise clipper
if resistor type spark plugs (such as those
made by Autolite) are employed. However, a
noise clipper is helpful if not absolutely necesmon
511
www.americanradiohistory.com
510
25 -10
Rotary
Beams
Indication of Direction
The most satisfactory method for indicating
the direction of transmission of a rotatable
array is that which uses Selsyns or synchros
for the transmission of the data from the rotating structure to the indicating pointer at the
operating position. A number of synchros and
Selsyns of various types are available on the
surplus market. Some of them are designed for
operation on 115 volts at 60 cycles, some are
designed for operation on 60 cycles but at a
lowered voltage, and some are designed for
operation from 400 -cycle or 800 -cycle energy.
This latter type of high- frequency synchro is
the most generally available type, and the
high- frequency units are smaller and lighter
than the 60 -cycle units. Since the indicating
synchro must deliver an almost negligible
amount of power to the pointer which it drives,
the high- frequency types will operate quite
satisfactorily from 60-cycle power if the voltage on them is reduced to somewhere between
6.3 and 20 volts. In the case of many of the
units available, a connection sheet is provided
along with a recommendation in regard to the
operating voltage when they are run on 60 cycles. In any event the operating voltage should
be held as low as it may be and still give satisfactory transmission of data from the antenna to the operating position. Certainly it should
not be necessary to run such a voltage on the
units that they become overheated.
A suitable Selsyn indicating system is shown
in figure 21.
Systems using a potentiometer capable of
continuous rotation and a milliammeter, along
with a battery or other source of direct current,
may also be used for the indication of direction. A commercially-available potentiometer
(Ohmite RB -2) may be used in conjunction
with a 0 -1 d -c milliammeter having a hand calibrated scale for direction indication.
25 -11
traps exert a minimum influence upon the element and it resonates at a frequency determined
by the electrical length of the configuration,
plus a slight degree of loading contributed by
the traps. At some higher frequency (generally
about 1.5 times the lowest operating frequency)
the outer set of traps are in a parallel resonant
condition, placing a high impedance between
the element and the tips beyond the traps.
Thus, the element resonates at a frequency
1.5 times higher than that determined by the
overall length of the element. As the frequency
of operation is raised to approximately 2.0
times the lowest operating frequency, the inner
set of traps become resonant, effectively disconnecting a larger portion of the element from
the driven section. The length of the center
section is resonant at the highest frequency
of operation. The center section, plus the two
adjacent inner sections are resonant at the
intermediate frequency of operation, and the
complete element is resonant at the lowest
frequency of operation.
The efficiency of such a system is determined by the accuracy of tuning of both the
element sections and the isolating traps. In
addition the combined dielectric losses of the
traps affect the overall antenna efficiency.
As with all multi -purpose devices, some compromise between operating convenience and
efficiency must be made with antennas designed
to operate over more than one narrow band of
frequencies. It is a tribute to the designers of
the better multi -band beams that they perform
as well as they do, taking into account the
theoretical difficulties that must be overcome.
r'
ISOLATING TRAPS -
"Three -Band" Beams
popular form of beam antenna introduced
during the past few years is the so- called
three -band beam. An array of this type is designed to operate on three adjacent amateur
bands, such as the ten, fifteen, and twenty
meter group. The principle of operation of
this form of antenna is to employ parallel
tuned circuits placed at critical positions in
the elements of the beam which serve to electrically connect and disconnect the outer sections of the elements as the frequency of excitation of the antenna is changed. A typical
'three -band" element is shown in figure 22.
At the lowest operating frequency, the tuned
it
A
FEED POINT
RESO NANT-
T
AT NIGNEST FREQUENCY
RESONANT
2
AT
INTERMEDIATE FREQUENCY
RESONANT
;
AT LOWEST FREQUENCY
Figure 22
TRAP -TYPE "THREE BAND"
ELEMENT
Isolating traps permit dipole to be
self-resonant at three widely different
www.americanradiohistory.com
frequencies.
HANDBOOK
-__
Antenna Control
Systems
ANTENNA ROTATOR
CONTROL BOX
r
S.P D.T.
RELAY
509
SOCKET
PLUG
SOCKET
1
e
1
PLUG
R
IS-CONTACT JONES PLUGS
SYNCHRO.
GENERATOR
L
r
D.P.D.T. TOGGLE SWITCH
1
fin
SOCKETS
ROTARY BEAM CONTROL
TO PROP MOTOR
INDICATOR
J
1
SYNCHRO.
PILOT
LIGHT
ITO 115 V.A.C.
i
TOGGLE
SWITCH
SOCKET
j
PLUG
SOCKET
L
PLUG
J
DIRECTION INDICATOR
Figure
21
SCHEMATIC OF A COMPLETE ANTENNA CONTROL SYSTEM
ating position as possible. However, on a particular installation the positions of the current
minimums on the transmission line near the
transmitter may be checked with the array in
the air, and then the array may be lowered to
ascertain whether or not the positions of these
points have moved. If they have not, and in
most cases if the feeder line is strung out back
and forth well above ground as the antenna is
lowered they will not change, the positions of
the last few toward the antenna itself may be
determined. Then the calculation of the matching quarter-wave section may be made, the section installed, the standing -wave ratio again
checked, and the antenna re- installed in its
final location.
25 -9
Antenna Rotation Systems
Structures for the rotation of antenna arrays
may be divided into two general classes: the
rotating mast and the rotating platform. The
rotating mast is especially suitable where the
transmitting equipment is installed in the garage or some structure away from the main
house. Such an installation is shown in figure
19. A very satisfactory rotation mechanism is
obtained by the use of a large steering wheel
located on the bottom pipe of the rotating mast,
with the thrust bearing for the structure located above the roof.
If the rotating mast is located a distance
from the operating position, a system of pulleys and drive rope may be used to turn the antenna, or a slow speed electric motor may be
employed.
The rotating platform system is best if a
tower or telephone pole is to be used for antenna support. A number of excellent rotating
platform devices are available on the market
for varying prices. The larger and more expensive rotating devices are suitable for the rdtaof a rather sizeable array for the 14-Mc. band
while the smaller structures, such as those
designed for rotating a TV antenna are designed for less load and should be used only with
a 28 -Mc. or 50 -Mc. array. Most common practice is to install the rotating device atop a
platform built at the top of a telephone pole
or on the top of a lattice mast of sizeable cross
section so that the mast will be self- supporting and capable of withstanding the torque imposed upon it by the rotating platform.
A heavy duty TV rotator may be employed
for rotation of 6 and 10 -meter arrays. Fifteen
and twenty meter arrays should use rotators
designed for amateur use such as the Cornell Dubilier HAM -I unit shown in figure 20.
www.americanradiohistory.com
Rotary
508
T H E
B e a m s
R A D
has been determined previously, and the
Antennascope control is turned for a null
readingon the meter of the Antennascope.
The impedance presented to the Antennascope by the matching device may be
read directly on the calibrated dial of
the Antennascope.
3. Adjustments should be made to the
matching device to present the desired
impedance transformation to the Antennascope. If a folded dipole is used as the
driven element, the transformation ratio
of the dipole must be varied as explained
previously in this chapter to provide a
more exact match. If a T -match or gamma
match system is used, the length of the
matching rod may be changed to effect
a proper match. If the Antennascope ohmic reading is lower than the desired reading, the length of the matching rod should
be increased. If the Antennascope reading is higher than the desired reading,
the length of the matching rod should be
decreased. After each change in length
of the matching rod, the series capacitor
in the matching system should be reresonated for best null on the meter of
the Antennascope.
Raising and
Lowering
the Array
P'
4041,,
A practical problem always present when tuning up and matching
an array is the physical location
of the structure. If the array is
atop the mast it is inaccessible for adjustment,
and if it is located on stepladders where it can
be adjusted easily it cannot be rotated. One
encouraging factor in this situation is the fact
that experience has shown that if the array is
placed 8 or 10 feet above ground on some stepladders for the preliminary tuning process, the
raising of the system to its full height will not
produce a serious change in the adjustments.
So it is usually possible to make preliminary
adjustments with the system located slightly
greater than head height above ground, and
then to raise the antenna to a position where
it may be rotated for final adjustments. If the
position of the sliding sections as determined
near the ground is marked so that the adjustments will not be lost, the array may be raised
to rotatable height and the fastening clamps
left loose enough so that the elements may be
slid in by means of a long bamboo pole. After
a series of trials a satisfactory set of lengths
can be obtained. But the end results usually
come so close to the figures given in figure 5
that a subsequent array is usually cut to the
dimensions given and installed as -is.
The matching process does not require rotation, but it does require that the antenna
proper be located at as nearly its normal oper-
Figure 20
HEAVY DUTY ROTATOR SUITABLE
FOR AMATEUR BEAMS
The new Cornell -Dubilier type HAM -1
rotor has extra heavy motor and gearing system to withstand weight and
inertia of amateur array under the
buffeting of heavy winds. Steel spur
gears and rotor lock prevent "pin wheeling" of antenna.
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I
O
Tuning the Array
HANDBOOK
a substantially separate process as just described. Alter
the tuning operation is complete, the resonant
frequency of the driven element of the antenna
should be checked, directly at the center of
the driven element if practicable, with a grid dip meter. It is important that the resonant frequency of the antenna be at the center of the
frequency band to be covered. If the resonant
frequency is found to be much different from
the desired frequency, the length of the driven
element of the array should be altered until
this condition exists. A relatively small change
in the length of the driven element will have
only a second order effect on the tuning of the
parasitic elements of the array. Hence, a moderate change in the length of the driven element may be made without repeating the tuning
process for the parasitic elements.
When the resonant frequency of the antenna
system is correct, the antenna transmission
line, with impedance -matching device or network between the line and antenna feed point,
is then attached to the array and coupled to a
low -power exciter unit or transmitter. Then,
preferably, a standing -wave meter is connected
in series with the antenna transmission line
at a point relatively much more close to the
transmitter than to the antenna. However, for
best indication there should be 10 to 15 feet
of line between the transmitter and the standing -wave meter. If a standing -wave meter is
not available the standing -wave ratio may be
checked approximately by means of a neon
lamp or a short fluorescent tube if twin transmission line is being used, or it may be check-
cess of tuning the array is made
ed with a thermomilliammeter and a loop, a
neon lamp, or an r -f ammeter and a pair of
clips spaced a fixed distance for clipping onto
one wire of a two -wire open line.
If the standing -wave ratio is below 1.5 to 1
Figure
507
it is satisfactory to leave the installation as
it is. If the ratio is greater than this range it
will be best when twin line or coaxial line is
being used, and advisable with open -wire line,
to attempt to decrease the s.w.r.
It must be remembered that no adjustments
made at the transmitter end of the transmission
line will alter the SWR on the line. All adjustments to better the SWR must be made at the
antenna end of the line and to the device which
performs the impedance transformation necessary to match the characteristic impedance of
the antenna to that of the transmission line.
Before any adjustments to the matching system are made, the resonant frequency of the
driven element must be ascertained, as explained previously. If all adjustments to correct impedance mismatch are made at this frequency, the problem of reactance termination
of the transmission line is eliminated, greatly
simplifying the problem. The following steps
should be taken to adjust the impedance transformation:
1. The output impedance of the matching
device should be measured. An Antenna scope and a grid -dip oscillator are required for this step. The Antennascope
is connected to the output terminals of
the matching device. If the driven element
is a folded dipole, the Antennascope
connects directly to the split section of
the dipole. If a gamma match or T -match
are used, the Antennascope connects to
the transmission -line end of the device.
If a Q- section is used, the Antennascope
connects to the bottom end of the section. The grid -dip oscillator is coupled
to the input terminals of the Antennascope as shown in figure 18.
2. The grid-dip oscillator is tuned to the
resonant frequency of the antenna, which
19
ALL -PIPE ROTATING MAST STRUCTURE FOR ROOF INSTALLATION
An installation suitable for a building with a pitched roof is shown at (A). At (B) is shown a
similar installation for a flat or shed roof. The arrangement as shown is strong enough to support a lightweight 3- element 28 -Mc. array and a light 3- element 50 -Mc. array above the 28 -Mc.
array on the end of a 4 -foot length of I/2-inch pipe.
The lengths of pipe shown were chosen so that when the system is in the lowered position one
can stand on a household ladder and put the beam in position atop the rotating pipe. The lengths
may safely be revised upward somewhat if the array is of a particularly lightweight design with
low wind resistance.
Just before the mast is installed It is a good idea to give the rotating pipe a good smearing of
cup grease or waterproof pump grease. To get the lip of the top of the stationary section of 1 %4inch pipe to project above the flange plate, it will be necessary to have a plumbing shop cut a
slightly deeper thread inside the flange plate, as well as cutting an unusually long thread on
the end of the 1%-inch pipe. It is relatively easy to waterproof this assembly through the roof
since the I 1/4-inch pipe is stationary at all times. Be sure to use pipe compound on all the joints
and then really tighten these joints with a pair of pipe wrenches.
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506
Rotary
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www.americanradiohistory.com
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s
HANDBOOK
Tuning the Array
driven onto a wooden dowel, as shown in
figure 178. The element may then be mounted
upon an aluminum support plate by means of
four ceramic insulators. Metal based insulators,
such as the Johnson 135 -67 are recommended,
since the all- ceramic types may break at the
mounting holes when the array is subjected
to heavy winds.
25 -8
Tuning the Array
Although satisfactory results may be obtained by pre -cutting the antenna array to the
dimensions given earlier in this chapter, the
occasion might arise when it is desired to
make a check on the operation of the antenna
before calling the job complete.
The process of tuning an array may fairly
satisfactorily be divided into two more or less
distinct steps: the actual tuning of the array
for best front -to-back ratio or for maximum forward gain, and the project of obtaining the
best possible impedance match between the
antenna transmission line and the feed point
of the array.
The actual tuning of the array
for best front -to -back ratio or
maximum forward gain may best
be accomplished with the aid of a low -power
transmitter feeding a dipole antenna (polarized
the same as the array being tuned) at least
four or five wavelengths away from the antenna being tuned and located at the same elevation as that of the antenna under test. A calibrated field- strength meter of the remote -indicating type is then coupled to the feed point
of the antenna array being tuned. The transmissions from the portable transmitter should
be made as short as possible and the call sign
of the station making the test should be transmitted at least every ten minutes.
It is, of course, possible to tune an array
with the receiver connected to it and with a
station a mile or two away making transmissions on your request. But this method is more
cumbersome and is not likely to give complete
satisfaction. It is also possible to carry out
the tuning process with the transmitter connected to the array and with the field- strength
meter connected to the remote dipole antenna.
In this event the indicating instrument of the
remote- indicating field - strength meter should
be visible from the position where the elements
are being tuned. However, when the array is
being tuned with the transmitter connected to
it there is always the problem of making continual adjustments to the transmitter so that a
constant amount of power will be fed to the
array under test. Also, if you use this system,
Tuning the
Array Proper
505
10 watts of power is
usually sufficient) and make sure that the antenna transmission line is effectively grounded
as far as d -c plate voltage is concerned. The
use of the method described in the previous
paragraph of course eliminates these problems.
One satisfactory method for tuning the array
proper, assuming that it is a system with several parasitic elements, is to set the directors
to the dimensions given in figure 5 and then
to adjust the reflector for maximum forward
signal. Then the first director should be varied
in length until maximum forward signal is obtained, and so on if additional directors are
used. Then the array may be reversed in direction and the reflector adjusted for best front to -back ratio. Subsequent small adjustments
may then be made in both the directors and the
reflector for best forward signal with a reasonable ratio of front -to -back signal. The adjustments in the directors and the reflector will
be found to be interdependent to a certain degree, but if small adjustments are made after
the preliminary tuning process a satisfactory
set of adjustments for maximum performance
will be obtained. It is usually best to make
the end sections of the elements smaller in
diameter so that they will slip inside the larger
tubing sections. The smaller sliding sections
may be clamped inside the larger main sections.
In making the adjustments described, it is
best to have the rectifying element of the remote- indicating field- strength meter directly
at the feed point of the array, with a resistor
at the feed point of the estimated value of
feed -point impedance for the array.
use very low power (5 or
Matching to the
Antenna Transmission Line
The problem of matching the
impedance of the antenna
transmission line to the array
is much simplified if the pro-
DRIVEN ELEMENT
ANTENNASCOK
RESONATING
CAPACITOR
RIDOIR MITER
Figure 18
ADJUSTMENT OF GAMMA MATCH BY USE
OF
ANTENNASCOPE
AND
GRID -DIP
METER
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THE
Rotary Beams
504
ELEMENT CLAMP
2 PIECES
LINE OF
ELEMENTS
J- ALUMINUM
APPROX.
S'
-
PLATE
X
LINE
BOOM, MADE
OF SECTIONS
OF STEEL TV
MAST OR OF
OF
ELEMENT
ALUMINUM
IRRIGATION
TUBING
12"
-
BOOM CLAMP
2 PIECES
O
PLATE WITH U- BOLTS,
REO'O) OR MUFFLER CLAMPS.
ELEMENT HELD
(2
U- BOLT
TO
LINE
\
SHIM JOINT WITH THIN
STRIPS OF ALUMINUM
IF NECESSARY
h
ADJUSTABLE
J2'
OF BOOM
RADIATOR
OXEN -YORE CLAMP
HOSE CLAMP
CENTER S_ECJI ON
LADDER
SLIT CENTER SECTION TUBE
SAT EACH
TIP
RADIO
ADJUSTABLE
END.
TIP
TYPICAL ELEMENT
O
Figure
16
ELEMENT "PLUMBER'S
DELIGHT" ANTENNA ARRAY
All-metal con figuration permits rugged,
light assembly. Joints are made with
U -bolts and metal plates for maximum
3-
ELEMENT HELD TO 2%4
BY 2 TV- TYPE U-BOLTS
BOLTED TO LADDER BY
2 PIECES OF ANGLE IRON STOCK
2 X
rigidity.
Figure
It is characteristic of
the conventional type of
multi -element parasitic
array such as discussed previously and outlined that the centers of all the elements are at
zero r-f potential with respect to ground. It is
therefore possible to use a metallic structure
without insulators for supporting the various
elements of the array. A typical three element
array of this type is shown in figure 16. In this
particular array, U-bolts and metal plates have
been employed to fasten the elements to the
boom. The elements are made of telescoping
sections of aluminum tubing. The tips of the
inner sections of tubing are split, and a tubing
clamp is slipped over the joint, as shown in
the drawing. Before assembly of the joint, the
mating pieces of aluminum are given a thin
coat of Penetrox-A compound. (This anti oxidizing paste is manufactured by l3urndy Co.,
Norwalk, Conn. and is distributed by the
General Electric Supply Co.) When the tubes
are telesooped and the clamp is tightened, an
air -tight seal is produced, reducing corrosion
"Plumber's Delight"
Construction
to a minimum.
The boom of the parasitic array may be made
from two or three sections of steel TV mast,
or it may be made of a single section of aluminum irrigation pipe. This pipe is made by
Reynolds Aluminum Co., and others, and may
often be purchased via the Sears, Roebuck Co.
mail -order department. Three inch pipe may be
17
(A) OXEN -YOKE CLAMP IS DESIGNED FOR
ALL METAL ASSEMBLY
(B) ALTERNATIVE WOODEN SUPPORTING
ARRANGEMENT
A
wooden ladder may be used to support
70 or 15 meter array.
a
used for the 10 and 15 meter antennas, and the
huskier four inch pipe should be used for a
20 meter beam.
Automobile muffler clamps can often be
to affix the elements to the support
plates. Larger clamps of this type will fasten
the plates to the boom. In most cases, the
muffler clamps are untreated, and they should
be given one or two coats of rust -proof paint
to protect them from inclement weather. All
bolts, nuts, and washers used in the assembly
of the array should be of the plated variety to
reduce corrosion and rust.
An alternative assembly is to employ the
`Oxen Yoke" type of clamps, shown in figure
17. These light- weight aluminum fittings are
obtainable from the Continental Electronics
and Sound Co., Dayton, 27, Ohio, and are
available in a wide range of sizes.
If it is desired to use a split driven element
for a balanced feed system, it is necessary to
insulate the element from the supporting structure of the antenna. The element should be
severed at the center, and the two halves
used
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Bi- directional
HANDBOOK
Arrays
FLAT -TOP BEAM
STUB
503
FOR
ROTATABLE ARRAY
GAIN
OPON -WIRE
TO B DB
LINE
O
"TWO OVER TWO OVER TWO*
TYPE OF ARRAY
Figure
GAIN
15
TOTAL NUMBER
OF
TWO GENERAL TYPES
OF BI- DIRECTIONAL
ARRAYS
Average gain figures are given for both the flat -top beam
type of array and for the
broadside- eel¡neor array with
different numbers of elements.
I.B
De
5.0
DB
ELEMENTS
z
7.5 DB
9.0
DB
10.0 DB
RAD!AL LOAD
45A
BEARING
D.
FEEDERS
N12 WIRE
SPACED
10
2'
GUY WIRES
ROPES TO
!NG POSITION
A.-THRUST
BEARING
availability of certain types of constructional
materials. But in any event be sure that sound
mechanical engineering principles are used in
the design of the supporting structure. There
are few things quite as discouraging as the
picking up of pieces, repairing of the roof, etc.,
when a newly constructed rotary comes down
in the first strong wind. If the principles of
mechanical engineering are understood it is
wise to calculate the loads and torques which
will exist in the various members of the structure with the highest wind velocity which may
be expected in the locality of the installation.
If this is not possible it will usually be worth
the time and effort to look up a friend who
understands these principles.
Radiating
Elements
One thing more or
less standard
about the construction of rotatable
antenna arrays is the use of durai
tubing for the self- supporting elements. Other
materials may be used but an alloy known as
24ST has proven over a period of time to be
quite satisfactory. Copper tubing is too heavy
for a given strength, and steel tubing, unless
copper plated, is likely to add an undesirably
large loss resistance to the array. Also, steel
tubing, even when plated, is not likely to
withstand salt atmosphere such as encountered
along the seashore for a satisfactory period
of time. Do not use a soft aluminum alloy for
the elements unless they will be quite short;
24ST is a hard alloy and is best although there
are several other alloys ending in "ST" which
will be found to be satisfactory. Do not use
an alloy ending in "SO" or "S" in a position
in the array where structural strength is important, since these letters designate a metal
which has not been heat treated for strength
and rigidity. However, these softer alloys, and
aluminum electrical conduit, may be used for
short radiating elements such as would be
used for the 50 -Mc. band or as interconnecting
conductors in a stacked array.
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502
Rotary
T
Beams
H E
R A D I O
O
"LAZY H" WITH REFLECTOR
GAIN APPROX. II De
BROADSIDE HALF -WAVES
WITH REFLECTORS
GAIN APPROX.
7
De
Figure 14
BROADSIDE ARRAYS
WITH PARASITIC
REFLECTORS
of the arrays illustrated will be greater than the values given due
to concentration of the radiated signal at the lower elevation ongles.
The apparent gain
TWO OVER TWO OVER TWO
WITH REFLECTORS
GAIN APPROX.
II.5
De
If six or more elements are used in the type
of array shown in figure 15B no matching section will be required between the antenna trans-
tuning will apply. However, the factor that a
bi- directional array need be rotated through an
angle of less than 180° should be considered
mission line and the feed point of the antenna.
When only four elements are used the antenna
is the familiar "lazy H" and a quarter -wave
stub should be used for feeding from the antenna transmission line to the feed point of
the antenna system.
If desired, and if mechanical considerations
permit, the gain of the arrays shown in figure
15B may be increased by 3 db by placing a
half-wave reflector behind each of the elements at a spacing of one -quarter wave. The
array then becomes essentially the same as
in this connection.
that shown in figure 14C and the same considerations in regard to reflector spacing and
25-7
Construction of
Rotatable Arrays
A considerable amount of ingenuity may be
exercised in the construction of the supporting
structure for a rotatable array. Every person
has his own ideas as to the best method of
construction. Often the most practicable method of construction will be dictated by the
HANDBOOK
Driven Arrays
linear system which will give approximately
the same gain as the system of figure 13A, but
which requires less boom length and greater
total element length. Figure 13C illustrates
the familiar lazy -H with driven reflectors (or
directors, depending upon the point of view)
in a combination which will show wide bandwidth with a considerable amount of forward
gain and good front -to -back ratio over the entire frequency coverage.
Three practicable
types of unidirectional
stacked broadside arrays are shown in figure 14. The first type,
shown at figure 14A, is the simple "lazy H"
type of antenna with parasitic reflectors for
each element. (B) shows a simpler antenna array with a pair of folded dipoles spaced one half wave vertically, operating with reflectors.
In figure 14C is shown a more complex array
with six half waves and six reflectors which
will give a very worthwhile amount of gain.
In all three of the antenna arrays shown the
spacing between the driven elements and the
reflectors has been shown as one -quarter wavelength. This has been done to eliminate the
requirement for tuning of the reflector, as a
result of the fact that a half -wave element
spaced exactly one - quarter wave from a driven
element will make a unidirectional array when
both elements are the same length. Using this
procedure will give a gain of 3 db with the reflectors over the gain without the reflectors,
with only a moderate decrease in the radiation
resistance of the driven element. Actually,
the radiation resistance of a half-wave dipole
goes down from 73 ohms to 60 ohms when an
identical half -wave element is placed one quarter wave behind it.
A very slight increase in gain for the entire
array (about 1 db) may be obtained at the expense of lowered radiation resistance, the necessity for tuning the reflectors, and decreased
bandwidth by placing the reflectors 0.15 wavelength behind the driven elements and making
them somewhat longer than the driven elements.
The radiation resistance of each element will
drop approximately to one -half the value obtained with untunedhalf-wave reflectors spaced
one -quarter wave behind the driven elements.
Antenna arrays of the type shown in figure
14 require the use of some sort of lattice work
for the supporting structure since the arrays
occupy appreciable distance in space in all
three planes.
Unidirectional Stacked
Broadside Arrays
The requirements for the feed
systems for antenna arrays of
the type shown in figure 14 are less critical
than those for the close- spaced parasitic arrays shown in the previous section. This is a
Feed Methods
501
natural result of the fact that a larger number
of the radiating elements are directly fed with
energy, and of the fact that the effective radiation resistance of each of the driven elements
of the array is much higher than the feed-point
resistance of a parasitic array. As a consequence of this fact, arrays of the type shown
in figure 14 can be expected to cover a somewhat greater frequency band for a specified
value of standing -wave ratio than the parasitic
type of array.
In most cases a simple open -wire line may
be coupled to the feed point of the array without any matching system. The standing -wave
ratio with such a system of feed will often be
less than 2 -to -1. However, if a more accurate
match between the antenna transmission line
and the array is desired a conventional quarter -wave stub, or a quarter-wave matching
transformer of appropriate impedance, may be
used to obtain a low standing-wave ratio.
25 -6
Bi- Directional
Rotatable Arrays
The bi- directional type of array is sometimes
used on the 28 -Mc. and 50 -Mc. bands where
signals are likely to be coming from only one
general direction at a time. Hence the sacrifice of discrimination against signals arriving
from the opposite direction is likely to be of
little disadvantage. Figure 15 shows two general types of bi- directional arrays. The flattop beam, which has been described in detail
earlier, is well adapted to installation atop a
rotating structure. When self-supporting elements are used in the flat -top beam the problem of losses due to insulators at the ends of
the elements is somewhat reduced. With a
single -section flat -top beam a gain of approximately 4 db can be expected, and with two
sections a gain of approximately 6 db can be
obtained.
Another type of bi- directional array which
has seen less use than it deserves is shown
in figure 15B. This type of antenna system has
a relatively broad azimuth or horizontal beam,
being capable of receiving signals with little
diminution in strength over approximately 40 °,
but it has a quite sharp elevation pattern since
substantially all radiation is concentrated at
the lower angles of radiation if more than a
total of four elements is used in the antenna
system. Figure 15B gives the approximate gain
over a half -wave dipole at the height of the
center of the array which can be expected. Also shown in this figure is a type of "rotating
mast" structure which is well suited to rotation of this type of array.
www.americanradiohistory.com
-
500
THE
Rotary Beams
__Ii/î/
\_----1
10111111---M1111
11---1111
-III
M1111 ro
1M
------MM
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En
D
111
11
1111
11NI
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Ill
5
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K
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IM1111111111111111111111 11111
4
DIRECTIONAL
N
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110
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50
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20
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10
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OGAIN ABOUT e De
FEED LINE
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DIRECTIONAL
e
SWR
Figure
RADIO
GAIN ABOUT e DB
12
SHORTED STUB LENGTH AND POSITION
CHART
From the standing wave ratio and current or
voltage null position it is possible to determine the theoretically correct length and
position of a shorted stub. In actual praca slight discrepancy usually will be
found between the theoretical and the experimentally optimized dimensions; therefore
it may be necessary to "touch up" the dimensions after using the above data as a
FEED LINE
tice
d4
DIRECTIONAL
GAIN ABOUT
10
De
starting point.
has been decided upon for the stub, and also
to determine the SWR.
Stub adjustment becomes more critical as
the SWR increases, and under conditions of
high SWR the current and voltage nulls are
more sharply defined than the current and voltage maxima, or loops. Therefore, it is best to
locate either a current null or voltage null, depending upon whether a current indicating device or a voltage indicating device is used to
check the standing wave pattern.
The SWR is determined by means of a "directional coupler," or by noting the ratio of
Ema. to Emin or La. to Imin as read on an
indicating device.
It is assumed that the characteristic impedance of the section of line used as a stub is
the same as that of the transmission line proper. It is preferable to have the stub section
identical to the line physically as well as
electrically.
25 -5
FEED LINE
Figure 13
UNIDIRECTIONAL ALL -DRIVEN ARRAYS
A unidirectional all -driven end-fire array is
shown at (A). (B) shows an array with two
half waves in phase with driven reflectors.
A Lazy -H array with driven reflectors is
shown at (C). Note that the directivity is
through the elements with the greatest total
feed -line length in arrays such as shown at
(B) and (C).
Unidirectional
Driven Arrays
place of a parasitic array of similar
dimensions when greater frequency coverage
than is available with the yagi type is desired.
Figure 13B is a combination end -fire and cobe used in
Three types of unidirectional driven arrays
are illustrated in figure 13. The array shown
in figure 13A is an end -fire system which may
www.americanradiohistory.com
HANDBOOK
The
Gamma
-FLAT' LINE
SWR
TO
Match
499
RESONANT
SECTION
1.O
TRANSMITTER
SIMPLE
OR
CONVEX
MATCHING STUB
Figure
10
THE GAMMA MATCHING SYSTEM
See text for details of resonating capacitor
piing rings are 10 inches in diameter and are
usually constructed of % -inch copper tubing
supported one from the rotating structure and
one from the fixed structure by means of standoff insulators. The capacitor C in figure 9D is
adjusted, after the antenna has been tuned, for
minimum standing -wave ratio on the antenna
transmission line. The dimensions shown will
allow operation with either 14 -Mc. or 28 -Mc.
elements, with appropriate adjustment of the
capacitor C. The rings must of course be parallel and must lie in a plane normal to the axis
of rotation of the rotating structure.
The Gamma Match
The use of coaxial cable
to feed the driven element
of a yagi array is becoming increasingly popular. One reason for this increased popularity
lies in the fact that the TVI- reduction problem
is simplified when coaxial feed line is used
from the transmitter to the antenna system.
Radiation from the feed line is minimized when
coaxial cable is used, since the outer conductor of the line may be grounded at several
points throughout its length and since the intense field is entirely confined within the outer conductor of the coaxial cable. Other advantages of coaxial cable as the antenna feed
line lie in the fact that coaxial cable may be
run within the structure of a building without
danger, or the cable may be run underground
without disturbing its operation. Also, transmitting -type low -pass filters for 52 ohm impedance are more widely available and are less
expensive than equivalent filters for two-wire
line.
The gamma -match is illustrated in figure 10,
and may be looked upon as one -half of a Tmatch. One resonating capacitor is used,
placed in series with the gamma rod. The capacitor should have a capacity of 7 µµfd. per
meter of wavelength. For 15-meter operation
the capacitor should have a maximum capacity
of 105 µµfd. The length of the gamma rod determines the impedance transformation between
Figure 11
IMPEDANCE MATCHING WITH A CLOSED
STUB ON A TWO WIRE TRANSMISSION
LINE
the transmission line and the driven element
of the array, and the gamma capacitor tunes
out the inductance of the gamma rod. By adjustment of the length of the gamma rod, and
the setting of the gamma capacitor, the SWR
on the coaxial line may be brought to a very
low value at the chosen operating frequency.
The use of an Antennascope, described in the
Test Equipment chapter is recommended for
precise adjustment of the gamma match.
The Matching Stub
If
an
open-wire line
is
used to feed a low impedance radiator, a section of the transmission
line may be employed as a matching stub as
shown in figure 11. The matching stub can
transform any complex impedance to the characteristic impedance of the transmission line.
While it is possible to obtain a perfect match
and good performance with either an open stub
or a shorted one by observing appropriate dimensions, a shorted stub is much more readily
adjusted. Therefore, the following discussion
will be confined to the problem of using a
closed stub to match a low impedance load to
a high impedance transmission line.
If the transmission line is so elevated that
adjustment of a "fundamental" shorted stub
cannot be accomplished easily from the ground,
then the stub length may be increased by ex-
actly one or two electrical half wavelengths,
without appreciably affecting its operation.
While the correct position of the shorting
bar and the point of attachment of the stub to
the line can be determined entirely by experimental methods, the fact that the two adjustments are interdependent or interlocking makes
such a cut- and -try procedure a tedious one.
Much time can be saved by determining the approximate adjustments required by reference to
a chart such as figure 12 and using them as a
starter. Usually only a slight "touching up"
will produce a perfect match and flat line.
In order to utilize figure 12, it is first necessary to locate accurately a voltage node or
current node on the line in the vicinity that
www.americanradiohistory.com
498
R o
t
Beams
y
a r
T
H E
R A D
I
O
OA DIRECT FEED WITH
COAXIAL CABLE
52 a. COAXIAL CABLE
0 QUARTER -WAVE
TRANSFORMER FEED
75
A TWIN LINE
450 -000 A. LINE
Figure
© TRANSFORMER
MATCHING SYSTEM
20 MC. - 4 TURNS
ANT.
14 MC.
-
METHODS WHERE THE
DRIVEN ELEMENT MAY
BE BROKEN IN THE
CENTER
2" DIA., 2"
1
9
ALTERNATE FEED
LONG
TURN EACH SIDE
TURNS 2" DIA., 2" LONG
ANT. TAPPED 2 TURNS EACH SIDE
0
0 COUPLINGLINK
ROTARY
COILS 10DIAMETER
COIL SPACED
APPROX.
0.5"
C
1
TURN LINKS ARE PARALLEL
C IS 200 LUPD VARIABLE
450 -0000. LINE
These capacitors should be tuned for minimum
SWR on the transmission line. The adjustment
of these capacitors should be made at the same
time the correct setting of the T -match rods is
made as the two adjustments tend to be interlocking. The use of the standing wave meter
(described in Test Equipment chapter) is
recommended for making these adjustments to
the T- match.
Four methods of exciting
the driven element of a
parasitic array are shown
in figure 9. The system
shown at (A) has proven to be quite satisfactory in the case of an antenna-reflector two element array or in the case of a three-element
array with 0.2 to 0.25 wavelength spacing between the elements of the antenna system. The
feed -point impedance of the center of the driven
element is close enough to the characteristic
impedance of the 52 -ohm coaxial cable so that
Feed Systems Using
Driven Element
with Center Feed
a
the standing -wave ratio on the 52 -ohm coaxial
cable is less than 2-to-1.(B) shows an arrangement for feeding an array with a broken driven
element from an open -wire line with the aid of
a quarter -wave matching transformer. With 465 ohm line from the transmitter to the antenna
this system will give a close match to a 12ohm impedance at the center of the driven element. (C) shows an arrangement which uses an
untuned transformer with lumped inductance
for matching the transmission line to the center impedance of the driven element.
Rotary Link
Coupling
In many cases it is desirable to
be able to allow the antenna ar-
ray to rotate continuously without
regard to snarling of the feed line. If this is to
be done some sort of slip rings or rotary joint
must be made in the feed line. One relatively
simple method of allowing unrestrained rotation
of the antenna is to use the method of rotary
link coupling shown in figure 9D. The two cou-
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HANDBOOK
Matching Systems
497
L
frT-u7r.
OA
L-.j
L
DELTA MATCH
DIMENSIONS SHOWN GIVE
APPROX. MATCH TO SOOD
AIN - SPACED LINE
L
Figure 8
AVERAGE DIMENSIONS
FOR THE DELTA AND
MATCH
1416
L
L
"T"
C -T
DI3Dz
MATCH
n Dn 300
TWIN LINE
zoo
In many cases it will be desired to use the
folded -element or yoke matching system with
different sizes of conductors or different spacings than those shown in figure 7. Note, then,
that the impedance transformation ratio of
these types of matching systems is dependent
both upon the ratio of conductor diameters and
upon their spacing. The following equation
has been given by Roberts (11CA Review, June,
1947) for the determination of the impedance
transformation when using different diameters
in the two sections of a folded element:
Transformation ratio
=
(1 +
\
-
Z,
z
Z,
this equation Z, is the characteristic impedance of a line made up of the smaller of
the two conductor diameters spaced the center to- center distance of the two conductors in the
antenna, and Z, is the characteristic impedance of a line made up of two conductors the
size of the larger of the two. This assumes
that the feed line will be connected in series
with the smaller of the two conductors so that
an impedance step up of greater than four will
be obtained. If an impedance step up of less
than four is desired, the feed line is connected
in series with the larger of the two conductors
and Z, in the above equation becomes the impedance of a hypothetical line made up of the
larger of the two conductors and Z2 is made
up of the smaller. The folded v -h -f unipole is
an example where the transmission line is connected in series with the a r g e r of the two
conductors.
In
1
n
The conventional 3 -wire match to give an
impedance 'multiplication of 9 and the 5 -wire
match to give a ratio of approximately 25 are
shown in figures 7C and 7D. The 4 -wire match,
not shown, will give an impedance transformation ratio of approximately 16.
The Delta match and the
T -match are shown in figure
8. The delta match has been
largely superseded by the newer T- match, however both these systems can be adjusted to
give a low value of SWR on 50 to 600 -ohm balanced transmission lines. In the case of the
systems shown it will be necessary to make
adjustments in the tapping distance along the
driven radiator until minimum standing waves
on the antenna transmission line are obtained.
Since it is sometimes impracticable to eliminate completely the standing waves from the
antenna transmission line when using these
matching systems, it is common practice to
cut the feed line, after standing waves have
been reduced to a minimum, to a length which
will give satisfactory loading of the transmitter
over the desired frequency range of operation.
The inherent reactance of the T -match is
tuned out by the use of two identical resonating capacitors in series with each leg of the
T -rod. These capacitors should each have a
maximum capacity of 8 littfd. per meter of wavelength. Thus for 20 meters, each capacitor
should have a maximum capacity of at least
160 µµEd. For power up to a kilowatt, 1000
volt spacing of the capacitors is adequate.
The Delta Match
and T-Match
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496
Rotary
T
B ea m s
RADIATION
Di
Di-Da
FOR
O
R.4
D11
FOR
MATCH
I
5 1.s Rs'8.9
2
s',RrecD
R A D
Di- 1Dzo.
FOR
FOLDED -ELEMENT
H E
D2.z5iegAre=10.5
S1.5
DI1
roa
D22.2s1 RAD.
5
D
1-
5
FOR D
=
*12 WI RE
3-
roua
D= 1-
*12
S
2WIRE
FOR
16
=11
-
RAD.
D= 15
5-
14
18
.
w 12 WIR1E
roa D. 1S= 1-
a0
WIRE
POR
D.
WIRE
12 S
=24
I
1
RAD.
32
Figure 7
DATA FOR
FOLDED -ELEMENT
MATCHING SYSTEMS
In all normal applications of
31M12. APPROX. 25
5 -WIRE MATCH
the data given the main element as shown is the driven
element of a multi -element
parasitic array. Directors and
reflectors have not been
shown for the sake of clarity.
R RAD.
small strips of polystyrene which have been
drilled for both the main element and the small
wire and threaded on the main element.
The calculation of the
operating conditions of
folded - element
the
matching system and the yoke match, as shown
in figures 7A and 7B is relatively simple. A
selected group of operating. conditions has
been shown on the drawing of figure 7. In applying the system it is only necessary to multiply the ratio of feed to radiation resistance
(given in the figures to the right of the suggested operating dimensions in figure 7) by
the radiation resistance of the antenna system
to obtain the impedance of the cable to be
used in feeding the array. Approximate values
of radiation resistance for a number of commonly used parasitic -element arrays are given
The Folded -Element
Match Calculations
in figure 5.
As an example, suppose a 3- element array
with 0.15D -0.15R spacing between elements is
to be fed by m e an s of a 465 -ohm line constructed of no. 12 wire spaced 2 inches. The
approximate radiation resistance of such an
antenna array will be 20 ohms. Hence we need
a ratio of impedance step up of 23 to obtain
a match between the characteristic impedance
of the transmission line and the radiation resistance of the driven element of the antenna
array. Inspection of the ratios given in figure
shows that the fourth set of dimensions
given under figure 7B will give a 24 -to -1 step
up, which is sufficiently close. So it is merely
necessary to use a 1 -inch diameter driven element with a no.8 wire spaced on 1 inch centers
(% inch below the outside wall of the 1 -inch
tubing) below the 1 -inch element. The no. 8
wire is broken and a 2 -inch insulator placed
in the center. The feed line then carries from
this insulator down to the transmitter. The
center insulator should be supported rigidly
from the 1 -inch tube so that the spacing between the piece of tubing and the no. 8 wire
will be accurately maintained.
7
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HANDBOOK
H-0.2
A
Arrays
Stacked Yagi
495
-wi.-13.2 A --4
H 0 2 -+A
O.
2
A--+- O.
2
A-.F
O
2 7.-.1
501
F Mc
DIRECTIONAL
DIRECTIONAL
A
OGAIN ABOUT 12 DB
WITH 2 SECTIONS
I
FEEDER LINE
©
O
GAIN ABOUT IS DB
WITH 3 SECTIONS
AIN ABOUT
17 DR
Figure 6
STACKED YAGI ARRAYS
It is possible to attain a relatively large amount of gain over a limited bandwidth with stacked
yogi arrays. The two -section array at (A) will give a gain of about 12 db, while adding a third
section will bring the gain up to about 15 db. Adding two additional parasitic directors to each
section, as at (C) will bring the gain up to about 17 db.
higher where the additional section of tubing
may be supported below the main radiator element without undue difficulty. The yoke-match
is more satisfactory mechanically on the 28-
bands since it is only necessary to suspend a wire below the driven element proper. The wire may be spaced below
the self-supporting element by means of several
Mc. and 14 -Mc.
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494
TYPE
Rotary
DRIVEN ELEMENT
LENGTH
473
F(MC)
3-ELEMENT
3- ELEMENT
THE
Beams
REFLECTOR
LENGTH
IST DIRECTOR
LENGTH
SOI
445
F(MC)
.F (MC)
F)
741?-41C)
4-ELEMENT
01(1°Z)
- --
2ND DIRECTOR 390 DIRECTOR SPACING BETLENGTH
LENGTH
WEENELEMENTS
-IS
.2S -.2S
.2 -.2 -.2
S(1SO
.1C
5-ELEMENT
.IS
F(MC)
Figure
F4(L9C)
.2
-.2 -.2-.2
small amount of additional
gain may be obtained through
use of more than two parasitic
elements, at the expense of reduced feed -point
impedance and lessened bandwidth. One additional director will add about 1 db, and a second additional director (making a total of five
elements including the driven element) will
add s l i g ht l y less than one db more. In the
v -h -f range, where the additional elements may
be added without much difficulty, and where
required bandwidths are small, the use of more
than two parasitic elements is quite practicable.
A
Three Elements
Parasitic arrays
(yagis) may
stacked to provide additional gain in the same manner that
dipoles may be stacked. Thus if an array of
six dipoles would give a gain of 10 db. the
substitution of yagi arrays for each of the dipoles would add the gain of one yagi array to
the gain obtained with the dipoles. However,
the yagi arrays must be more widely spaced
than the dipoles to obtain this theoretical improvement. As an example, if six 5- element
yagi arrays having a gain of about 10 db were
substituted for the dipoles, with appropriate
increase in the spacing between the arrays,
the gain of the whole system would approach
the sum of the two gains, or 20 db. A group of
arrays of yagi antennas, with recommended
spacing and approximate gains, are illustrated in figure 6.
Stacking of
Yogi Arrays
25 -4
be
Feed Systems for
Parasitic (Yogi) Arrays
The table of figure 5 gives, in addition to
other information, the approximate radiation
resistance referred to the center of the driven
element of multi- element parasitic arrays. It is
obvious, from these low values of radiation
APPRO %.RADIATION
RESISTANCE
7.5
20
9.S
35
9.S
20
Io.o
IS
(A
I
5
DESIGN CHART FOR PARASITIC ARRAYS (DIMENSIONS
GIVEN
More Than
APPROX. GAIN
DO
RADIO
IN
FEET)
resistance, that especial care must be taken
in materials used and in the construction of
the elements of the array to insure that ohmic
losses in the conductors will not be an appreciable percentage of the radiation resistance.
It is also obvious that some method of iglpedance transformation must be used in many
cases to match the low radiation resistance
of these antenna arrays to the normal range of
characteristic impedance used for antenna
transmission lines.
A group of possible methods of impedance
matching is shown in figures 7, 8, 9 and 10.
All these methods have been used but certain
of them offer advantages over some of the
other methods. Generally speaking it is not
mechanically desirable to break the center of
the driven element of an array for feeding the
system. Breaking the driven element rules out
the practicability of building an all -metal or
"plumber's delight" type of array, and imposes mechanical limitations with any type of
construction. However, when continuous rotation is desired, an arrangement such as shown
in figure 9D utilizing a broken driven element
with a rotatable transformer for coupling from
the antenna transmission line to the driven
element has proven to be quite satisfactory.
In fact the method shown in figure 9D is probably the most practicable method of feeding
the driven element when continuous rotation
of the antenna array is required.
The feed systems shown in figure 7 will,
under normal conditions, show the lowest losses of any type of feed system since the currents flowing in the matching network are the
lowest of all the systems commonly used. The
"Folded Element" match shown in figure 7A
and the "Yoke" match shown in figure 7B are
the most satisfactory electrically of all standard feed methods. However, both methods require the extension of an additional conductor
out to the end of the driven element as a portion of the matching system. The folded -element match is best on the 50 -Mc. band and
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Parasitic Arrays
HANDBOOK
0.2 wavelength between elements becomes
possible. Four -element arrays are quite common on the 28 -Mc. and 50 -Mc. bands, and five
elements are sometimes used for increased
gain and discrimination. As the number of elements is increased the gain and front-to -back
ratio increases but the radiation resistance decreases and the bandwidth or frequency range
over which the antenna will operate without
reduction in effectiveness is decreased.
While the elements may consist
supported on a wood
framework, self-supporting elements of tubing are much to be preferred. The
latter type array is easier to construct, looks
better, is no more expensive, and avoids the
problem of getting sufficiently good insulation
at the ends of the elements. The voltages
reach such high values towards the ends of
the elements that losses will be excessive,
unless the insulation is excellent.
The elements may be fabricated of thin walled steel conduit, or hard drawn thin -walled
copper tubing, but durai tubing is much better.
Or, if you prefer, you may purchase tapered
copper-plated steel tubing elements designed
especially for the purpose. Kits are available
complete with rotating mechanism and direction
indicator, for those who desire to purchase
the whole system ready to put up.
Material for
Elements
of wire
The optimum spacing for a
two -element array is, as has
been mentioned be fore, approximately 0.11
wavelength for a director and 0.13 wavelength
for a reflector. However, when both a director
and a reflector are combined with the driven
element to make up a three-element array the
optimum spacing is established by the bandwidth which the antenna will be required to
cover. Wide spacing (of the order of 0.25 wavelength between elements) will result in greater
bandwidth for a specified maximum standing wave ratio on the antenna transmission line.
Smaller spacings may be used when boom
length is an important consideration, but for a
specified standing-wave ratio and forward gain
the frequency coverage will be smaller. Thus
the Q of the antenna system will be increased
as the spacing between the elements is decreased, resulting in smaller frequency coverage, and at the same time the feed -point impedance of the driven element will be decreased.
For broad -band coverage, such as the range
from 26.96 to 29.7 Mc. or from 50 to 54 Mc.,
0.2 wavelength spacing from the driven element to each of the parasitic elements is recElement Spacing
493
For narrower bandwidth, such as
would be adequate for the 14.0 to 14.4 Mc.
band or the 144 to 148 Mc. band, the radiator
to parasitic element spacing may be reduced
to 0.12 wavelength, while still maintaining
adequate array bandwidth for the amateur band
in question.
ommended.
Experience has shown that
it is practical to cut the
prarsitic elements of a
three -element parasitic array to a predetermined
length before the installation of such an antenna. A pre -tuned antenna such as this will
give good signal gain, adequate front -to -back
ratio, and good bandwidth factor. By carefully
tuning the array after it is in position the gain
may be increased by a fraction of a db, and
the front -to -back ratio by several db. However
the slight improvement in performance is usually not worth the effort expended in tuning
time.
The closer the lengths of the parasitic elements are to the resonant length of the driven
element, the lower will be the feed -point resistance of the driven element, and the smaller
will be the bandwidth of the array. Hence, for
wide frequency coverage the director should
be considerably shorter, and the reflector considerably longer than the driven element. For
example, the director should still be less than
a resonant half wave at the upper frequency
limit of the range wherein the antenna is to be
operated, and the reflector should still be long
enough to act as a reflector at the lower frequency limit. Another way of stating the same
thing is to say, in the case of an array to cover
a wide frequency range such as the amateur
range from 26.96 to 29.7 Mc. or the width of a
low -band TV channel, that the director should
be cut for the upper end of the band and
the reflector for the lower end of the band. In
the case of the 26.96 to 29.7 Mc. range this
means that the director should be about 8 per
cent shorter than the driven element and the
reflector should be about 8 per cent longer.
Such an antenna will show a relatively constant gain of about 6 db over its range of coverage, and the pattern will not reverse at any
point in the range.
Where the frequency range to be covered is
somewhat less, such as a high -band TV channel, the 14.0 to 14.4 Mc. amateur band, or the
lower half of the amateur 28 -Mc. phone band,
the reflector should be about 5 per cent longer
than the driven element, and the director about
5 per cent shorter. Such an antenna will perform well over its rated frequency band, will
not reverse its pattern over this band, and will
show a signal gain of 7 to 8 db. See figure 5
for design figures for 3-element arrays.
Length of the
Parasitic Elements
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492
THE
Rotary Beams
RADIO
wavelength may be employed for greater front to -back ratios, but the radiation resistance of
the array becomes quite low, the bandwidth
of the array becomes very narrow, and the tuning becomes quite critical. Thus the Q of the
antenna system will be increased as the spacing between the elements is decreased, and
smaller optimum f r e q u e n c y coverage will
result.
z
o
Element Lengths
When the parasitic element
of a two -element array is
used as a director, the following formulas may
be used to determine the lengths of the driven
element and the parasitic director, assuming
an element diameter -to- length ratio of 200 to
400:
476
Driven element length (feet)
-
=
=
FAlc.
Element spacing (feet)
Figure
=
l
11c.
4
FIVE ELEMENT 28 MC BEAM
ANTENNA AT W6SAI
Antenna boom is made of twenty foot
length of Sears, Roebuck Co. threeinch aluminum irrigation pipe. Spacing
between elements is five feet. Elements are made of twelve foot lengths
of 7/8 -inch aluminum tubing, with extension tips made of 3/4 -inch tubing.
Gamma
matching device, element
clamps, and 'Oxen Yoke" element -toboom clamps are made by Continental
Electronics
8 Sound Co., Dayton 27,
Ohio. Beam dimensions are taken from
figure
5.
o.IS
oz
0.2S
ELEMENT SPACING (X)
Figure 3
FRONT -TO -BACK RATIO AS A FUNCTION
OF ELEMENT SPACING FOR A TWO -ELEMENT PARASITIC ARRAY
Fmc.
450
Director length (feet)
0 I
(PARASITIC ELEMENT TUNED FOR MAXIMUM GAIN)
The effective bandwidth taken between the
1.5/1 standing wave points of an array cut to
the above dimensions is about 2.5% of the
operating frequency. This means that an array
pre -cut to a frequency of 14,150 kilocycles
would have a bandwidth of 350 kilocycles (plus
or minus 175 kilocycles of the center frequency), and therefore would be effective over the
whole 20 meter band. In like fashion, a 15
meter array should be pre -cut to 21,200 kilo-
cycles.
A beam designed for use on the 10 -meter
band would have an effective bandwidth of
some 700 kilocycles. Since the 10 -meter band
is 1700 kilocycles in width, the array should
either be cut to 28,500 kilocycles for operation in the low frequency portion of the band,
or to 29,200 kilocycles for operation in the
high frequency portion of the band. Operation
of the antenna outside the effective bandwidth
will increase the SWR on the transmission
line, and noticeably degrade both the gain and
front -to-back ratio performance. The height
above ground also influences the F/B ratio.
25 -3
The Three -Element Array
The three -element array using a director,
driven element, and reflector will exhibit as
much as 30 db front -to -back ratio and 20 db
front -to -side ratio for low -angle radiation. The
theoretical gain is about 9 db over a dipole in
free space. In actual practice, the array will
often show 7 to 10 db apparent gain over a
horizontal dipole placed the same height above
ground (at 28 and 14 Mc.).
The use of more than three elements is desirable when the length of the supporting structure is such that spacings of approximately
www.americanradiohistory.com
Parasitic Arrays
491
SO
45
k
40
r
35
30
t
..
.
zs
20
.....
»
Io
s
0
1
0.15
0 2
ais
ELEMENT SPACING (X)
Figure
0
1
0.15
0
2
ELEMENT SPACING (X)
1
GAIN VS ELEMENT SPACING FOR A TWO ELEMENT CLOSE- SPACED PARASITIC
BEAM ANTENNA WITH PARASITIC ELEMENT OPERATING AS A DIRECTOR OR
REFLECTOR
Such an antenna is capable of a signal gain
of 5 db over a dipole, with a front -to -back ratio
of 7 db to 15 db, depending upon the adjustment of the parasitic element. The parasitic
element may be used either as a director or
as a reflector.
The optimum spacing for a reflector in a
two -element array is approximately 0.13 wavelength and with optimum adjustment of the
length of the reflector a gain of approximately
5 db will be obtained, with a feed -point resistance of about 25 ohms.
If the parasitic element is to be used as a
director the optimum spacing between it and
the driven element is 0.11 wavelength. The
gain will theoretically be slightly greater than
with the optimum adjustment for a reflector
(about 5.5 db) and the radiation resistance
will be in the vicinity of 17 ohms.
The general characteristics of a two -element
parasitic array may be seen in figures 1, 2 and
3. The gain characteristics of a two -element
array when the parasitic element is used as a
director or as a reflector are shown. It can be
seen that the director provides a maximum of
5.3 db gain at a spacing of slightly greater
than 0.1 wavelength from the antenna. In the
interests of greatest power gain and size conservation, therefore, the choice of a parasitic
director would be wiser than the choice of a
parasitic reflector, although the gain difference between the two is small.
Figure 2 shows the relationship between
the element spacing and the radiation resist-
Figure
2
RADIATION RESISTANCE AS A FUNCTION
OF ELEMENT SPACING FOR A TWO -ELEMENT PARASITIC ARRAY
ance for the two element parasitic array for
both the reflector and the director case. Since
the optimum antenna-director spacing for maximum gain results in an antenna radiation resistance of about 17 ohms, and the optimum
antenna- reflector spacing for maximum gain
results in an antenna radiation resistance of
about 25 ohms, it may be of advantage in some
instances to choose the antenna with the higher radiation resistance, assuming other factors to be equal.
Figure 3 shows the front -to -back ratio for
the two element parasitic array for both the
reflector and director cases. To produce these
curves, the elements were tuned for maximum
gain of the array. Better front -to -back ratios
may be obtained at the expense of array gain,
if desired, but the general shape of the curves
remains the same. It can be readily observed
that operation of the parasitic element as a
reflector produces relatively poor front -toback ratios except when the element spacing
is greater than 0.15 wavelength. However, at
this element spacing, the gain of the array begins to suffer.
Since a radiation resistance of 17 ohms is
not unduly hard to match, it can be argued that
the best all- around performance may be obtained from a two element parasitic beam employing 0.11 element spacing, with the parasitic element tuned to operate as a director.
This antenna will provide a forward gain of
5.3 db, with a front -to -back ratio of 10 db, or
slightly greater. Closer spacing than 0.11
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CHAPTER TWENTY -FIVE
Rotary Beams
The rotatable antenna array has become almost standard equipment for operation on the
28 -Mc. and 50 -Mc. bands and is commonly used
on the 14-Mc. and 21 -Mc. bands and on those
frequencies above 144 Mc. The rotatable array
offers many advantages for both military and
amateur use. The directivity of the antenna
types commonly employed, particularly the
unidirectional arrays, offers a worthwhile reduction in interference from undesired directions. Also, the increase in the ratio of low angle radiation plus the theoretical gain of
such arrays results in a relatively large increase in both the transmitted signal and the
signal intensity from a station being received.
A significant advantage of a rotatable antenna array in the case of the normal station is
that a relatively small amount of space is required for erection of the antenna system. In
fact, one of the best types of installation uses
a single telephone pole with the rotating structure holding the antenna mounted atop the pole.
To obtain results in all azimuth directions
from fixed arrays comparable to the gain and
directivity of a single rotatable three- element
parasitic beam would require several acres of
surface.
There are two normal configurations of radiating elements which, when horizontally polarized, will contribute to obtaining a low angle
of radiation. These configurations are the end fire array and the broadside array. The con-
ventional three- or four -element rotary beam
may properly be called a unidirectional parasitic end-fire array, and is actually a type of
yagi array. The flat -top beam is a type of bidirectional end-lire array. The broadside type
of array is also quite effective in obtaining
low -angle radiation, and although widely used
in FM and TV broadcasting has seen little use
by amateur stations in rotatable arrays.
25 -1
Unidirectional
Parasitic End -Fire Arrays
(Yogi Type)
If a single parasitic element is placed on
one side of a driven dipole at a distance of
from 0.1 to 0.25 wavelength the parasitic element can be tuned to make the array substantially unidirectional.
This simple array is termed
parasitic beam.
25 -2
a
two element
The Two Element Beam
The two element parasitic beam provides
the greatest amount of gain per unit size of
any array commonly used by radio amateurs.
490
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HANDBOOK
Parasitic Arrays
VHF
489
DRILL HOLES THROUGH BOOM AND
PASS ELEMENTS THROUGH HOLES
BOOM LENGTH
DRIVEN ELEMENT
GAIN= 16.1
ELEMENT DIMENSIONS
REFLECTOR
DIRECTORS
144 MC.
145 MC.
146 MC.
147 MC.
41^
4ot
4°4'
404--
36
367
^
24
.
DIAM.
If
DB
2 METER BAND
LENGTH
ELEMENT
(DIA M. I /6 -)
=
36a
SPACING
FROM
DIPOLE
19
D1=
7
02= 14.5
DRIVEN ELEMENT
36.5
D3=
z2^
D4=
36
DS=
70.
De=
loz
De=
134
lee.
D9=
19e
D7=
e WIRE FOR 300 (1
MATCH.
*10 WIRE
MATCH
FOR
INSULATING
PLATE
4500
BLATTEN
TUBING
AT ENDS.
D10=230"
D11=242"
Figure
DESIGN
DIMENSIONS FOR
A
2-METER "LONG YAGI" ANTENNA
other hand, if a Yagi array of the same
approximate size and weight as another antenna type is built, it will provide a higher
order of power gain and directivity than that
of the other antenna.
The power gain of a Yagi antenna increases
directly with the physical length of the array.
The maximum practical length is entirely a
mechanical problem of physically supporting
the long series of director elements, although
when the array exceeds a few wavelengths in
length the element lengths, spacings, and
Q's become more and more critical. The effectiveness of the array depends upon a proper
combination of the mutual coupling loops
between adjacent directors and between the
first director and the driven element.
On the
Practically all work on Yagi antennas with
more than three or four elements has been on
an experimental, cut- and -try basis. Figure 19
19
provides dimensions for a typical Long Yagi
antenna for the 2 -meter VHF band. Note that
all directors have the same physical length.
If the long Yagi is designed so that the directors gradually decrease in length as they
progress from the dipole bandwidth will be
increased, and both side lobes and forward
gain will be reduced. One advantage gained
from staggered director length is that the
array can be shortened and lengthened by
adding or taking away directors without the
need for retuning the remaining group of parasitic elements. When all directors are the
same length, they must be all shortened en
masse as the array is lengthened, and viceversa when the array is shortened.
A full discussion of Long Yagi antennas,
including complete design and construction
information may be had in the VHF Handbook,
available through Radio Publications, Inc.,
Wilton., Conn.
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488
V -H -F
and
U -H
-F
THE
Antenn as
RADIO
WOODBLOCK
-BRASS TUBING
A
STUB
C
Figure
16
THE MOUNTING BLOCK FOR EACH SET
OF ELEMENTS
TRANSFORMER
-B
SHORTING BAR -C
These tubes are welded onto the center tube
of each group of three horizontal bracing tubes,
and are so located to support the horizontal dipole at its exact center. The dipoles are attached to the supporting rods by means of
small phenolic insulating blocks, as shown in
figure 16. The radiators are therefore insulated
from the screen reflector. The inner tips of
the radiators are held by small polystyrene
blocks for rigidity, and are cross connected to
each other by a transposed length of TV -type
400 ohm open wire line. The entire array is
fed at the point A -A, illustrated in figure 15.
The matching system for the beam is mounted
behind the reflector screen, and is shown in
figure 17. A quarter -wave transformer (B) drops
the relatively high impedance of the antenna
array to a suitable value for the low impedance balun (D). An adjustable matching stub
(C) and two variable capacitors (C, and C2)
are employed for impedance matching. The
two variable capacitors are mounted in a
I
C1&C2
=SO LUF
WATERTIGHT
COMPARTMENT
APPROX
r
BALUN- D
-- SNORTING BAR - D
COPPER TUBING
72 R COAX CABLE
Figure 17
THE MATCHING UNIT IN DETAIL FOR
THE PE1PL BEAM DESIGN, WHICH ALLOWS THE USE OF 72 -OHM COAX
watertight box, with the balun and matching
stubs entering the bottom and top of the box,
respectively.
The matching procedure is carried out by
the use of a standing wave meter (SWR bridge).
A few watts of power are fed to the array
through the SWR meter, and the setting of the
shorting stub on C and the setting of the two
variable capacitors are adjusted for lowest
SüR at the chosen operating frequency. The
capacity settings of the two variable capacitors should be equal. The final adjustment is
to set the shorting stub of the balun (D) to remove any residual reactance that might appear
on the transmission line. üith proper adjust-
Figure 18
HORIZONTAL RADIATION PATTERN OF
THE PE1PL ARRAY. THE FRONT -TOBACK RATIO IS ABOUT 28 db IN AMPLITUDE, AND THE FORWARD GAIN APPROXIMATELY 15 db.
ment, the VSWR of the array may be held to
less than 1.5 to 1 over a 2 megacycle range
of the 2 -meter band.
The horizontal radiation pattern of this array
is shown in figure 18.
a given power gain, the
Yagi antenna can be built
lighter, more compact, and
with less wind resistance than any other type.
Long Yogi
Antennas
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For
HANDBOOK
VHF
Parasitic Arrays
487
The ends of the folded dipoles are made in
following manner: Drive a length of dowel
into the short connecting lengths of aluminum
tubing. Then drill down the center of the dowel
the
with
a
clearance hole for the connecting screw.
Then shape the ends of the connecting pieces
to fit the sides of the element ends. After assembly the junctions may be dressed with a
file and sandpaper until a smooth fit is ob-
tained.
The mast used for supporting the array is a
30 -foot
spliced 2 by 2. A large discarded ball
bearing is used as the radial load bearing and
guy -wire termination. Enough of the upper-mast
corners were removed with a draw-knife to permit sliding the ball bearing down about 9 feet
from the top of the mast. The bearing then was
encircled by an assembly of three pieces of
dural ribbon to form a clamp, with ears for
tightening screws and attachment of the guy
wires. The bearing then was greased and covered with a piece of auto inner tube to serve
as protection from the weather. Another junk box bearing was used at the bottom of the mast
as a thrust bearing.
The main boom s were made from 34-inch
aluminum electrical conduit. Any size of small
tubing will serve for making the elements.
Note that the main boom is mounted at the balance center and not necessarily at the physical center. The pivot bolt in the offset head
should be tightened sufficiently that there will
be adequate friction to hold the array in position. Then an additional nut should be placed
on the pivot bolt as a lock.
In connecting the phasing sections between
the T- junction and the centers of the folded
dipoles, it is important that the center conductors of the phasing sections be connected
to the same side of the driven elements of the
antennas. In other words, when the antenna is
oriented for horizontal polarization and the
center of the coaxial phasing section goes to
the left side of the top antenna, the center
conductor of the other coaxial phasing section
should go to the left side of the bottom antenn a.
The "Screen Beam"
for 2 Meters
This highly effective ro-
tary array for the 144 Mc.
amateur band was designed by the staff of the Experimental Physics Laboratory, The Hague, Netherlands for
use at the 2 meter experimental station PEIPL.
The array consists of 10 half wave radiators
fed in phase, and arranged in two stacked rows
of five radiators. 0.2 wavelength behind this
plane of radiators is a reflector screen, measuring approximately 15' x 9' in size. The antenna provides a power gain of 15 db, and a
front to back ratio of approximately 28 db.
ALL JOINTS WELDED
Figure 15
DETAIL OF LAYOUT AND
OF BEAM ASSEMBLY
OF
DIMENSIONS
PEIPL
The 10 dipoles are fed in phase by means
of a length of balanced transmission line, a
quarter -wave matching transformer, and a balun. A 72 -ohm coaxial line couples the array
to the transmitter. A drawing of the array is
shown in figure 15.
The reflecting screen measures 14' 9" high
by 8' 4" wide, and is made of welded %" diameter steel tubing. Three steel reinforcing
bars are welded horizontally across the framework directly behind each pair of horizontal
dipoles. The intervening spaces are filled
with lengths of no. 12 enamel- coated copper
wire to complete the screen. The spacing between the wires is 2 ". Four cross braces are
welded to the corners of the frame for additional bracing, and a single vertical %" rod
runs up the middle of the frame. The complete,
welded frame is shown in figure 15. The no.
12 screening wires are run between 6-32 bolts
placed in holes drilled in each outside vertical member of the frame.
The antenna assembly is supported away
from the reflector screen by means of ten
lengths of % " steel tubing, each l' 3%4" long.
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486
V -H -F
and
U
-H -F
THE
Antennas
RADIO
Figure 14
THE EIGHTELEMENT 144 -MC. ARRAY IN A HORIZONTAL POSITION
appropriate cord. Hence, the operation is based
on the offset head sketched in figure 13. Although a wood mast has been used, the same
system may be used with a pipe mast.
The 40 -inch lengths of RG -59/U cable (electrically 3i4 wavelength) running from the center
of each folded dipole driven element to the
coaxial T- junction allow enough slack to permit free movement of the main boom when
changing polarity. Type RG -8 /U cable is run
from the T- junction to the operating position.
Measured standing -wave ratio was less than
2:1 over the 144 to 148 Mc. band, with the
lengths and spacings given in figure 13.
Construeion of
the Array
Most of the constructional
aspects of the antenna array
are
self- evident from figure
13. However, the pointers given in the following paragraphs will be of assistance to those
wishing to reproduce the array.
The drilling of holes for the small elements
should be done carefully on accurately marked
centers. A small angular error in the drilling
of these holes will result in a considerable
misalignment of the elements after the array is
assembled. The same consideration is true of
the filing out of the rounded notches in the
ends of the main boom for the fitting of the
two antenna booms.
Short lengths of wood dowel are used freely
in the construction of the array. The ends of
the small elements are plugged with an inch
or so of dowel, and the ends of the antenna
booms are similarly treated with larger discs
pressed into place.
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HANDBOOK
VHF
w- -,6
16
Parasitic Arrays
485
2ND
DIRECTOR
35 5"
I
___1ST
DIRECTOR
36"
RADIATOR
35"
REFLECTOR
40"
5
RING BOLT
ß_4B
60-
TO
FIT
ELEMENTS
--
O.D.
-
BOOM
RG -59 /U
EACH
MAIN BOOMS- S- APPROX.
BOOM
FILE END
CABLES
40 LONG
-B /U CABLE
' T. COAXIAL
FITTING
RG
TO
SHAPE ENDS OF SHORT PIECES
TO FIT CONTOUR
INSULATING ROO.
ENDS
CUT DOWN TO GO INTO TUBING
ABOU
ENDS OF TUBING
WOOD DOWELS INSIDE FOR STRENGTH
TERMINALS
-
AS SHOWN.
ANTENNA IS
HORIZONTALLY POLARIZED
PULL TO SWING MAIN BOOM
FOR VERTICAL POLARITY.
90
CONTROL CORDS
RG -13/U
TO RIG
' 2X2
ROTATABLE
CA
WOOD
MAST
RADIAL BEARING
CONSTRUCTIONAL
DRAWING
OF
AN
Figure 13
EIGHT- ELEMENT
quency range. Although polarization has been
loosely standardized in various areas of the
country, exceptions are frequent enough so
that it is desirable that the polarization of antenna radiation be easily changeable from horizontal to vertical.
The antenna illustrated has shown a signal
gain of about 11 db, representing a power gain
of about 13. Although the signal gain of the
TIPPABLE
144 -MC.
ARRAY
antenna is the same whether it is oriented for
vertical or horizontal polarization, the horizontal beam width is smaller when the antenna
is oriented for vertical polarization. Conversely, the vertical pattern is the sharper when
the antenna system is oriented for horizontal
polarization.
The changeover from one polarization to the
other is accomplished simply by pulling on the
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484
V -H -F
and
U -H
sistors in series. If 2 -watt resistors are employed, this termination also is suitable for
transmitter outputs of 10 watts or less. For
higher powers, however, resistors having greater dissipation with negligible reactance in the
upper v -h -f range are not readily available.
For powers up to several hundred watts a
suitable termination consists of a "lossy"
line consisting of stainless steel wire (corresponding to no. 24 or 26 B &S gauge) spaced 2
inches, which in turn is terminated by two
390 -ohm 2 -watt carbon resistors. The dissipative line should be at least 6 wavelengths
long.
FOR
1
D=22^
¡
METERS
D-A-9
A-23}
R-y
R-
REFLECTOR
40 LONG
MWÚS2-
BEND RADIUS
1B
-
FEED LINE
TNRU HOLE,
MIP-D
1X2 WELL-SEASONED
WOOD
GAIN 7.5 DB
(20. LONG
FOR
6/32
f
34 LONG
FOR 11- METERS)
SCREWS
TAIL
DIRECTOR
36" LONG
Y
1)
75
TV LINE
The rotary multi -element beam is undoubtedly the most popular type of v -h-f antenna in
use. In general, the design, assembly and tuning of these antennas follows a pattern similar
to the iarger types of rotary beam antennas
used on the lower frequency amateur bands.
The characteristics of these low frequency
beam antennas are discussed in the next chapter of this Handbook, and the information contained in that chapter applies in general to the
v -h -f beam antennas discussed herewith.
simplest v -h -f beam for
the beginner is the three -element Yagi array illustrated in
figure 12. Dimensions are
given for Yagis cut for the 2 -meter and ISmeter bands. The supporting boom for the Yagi
may be made from a smoothed piece of 1" x 2"
wood. The wood should be reasonably dry and
should be painted to prevent warpage from exposure to sun and rain. The director and reflector are cut from lengths of %" copper tubing, obtainable from any appliance store that
does service work on refrigerators. They should
be cut to length as noted in figure 12. The elements should then be given a coat of aluminum
paint. Two small holes are drilled at the center
of the reflector and director and these elements
are bolted to the wood boom by means of two
wood screws. These screws should be of
111
the plated, or rust -proof variety.
The driven element is made of a 78" length
of ia" copper tubing, the ends bent back upon
each other to form a folded dipole. If the tubing is packed with fine sand and the bending
points heated over a torch, no trouble will be
had in the bending process. If the tubing does
collapse when it is bent, the break may be reThe
Element Beam
Antenna
a
.il
WOOD BOOM
Beam Antennas
paired with
_
L
N3ULATING_BLbC5
Multi- Element V -H -F
A Simple Three
D
A-DRIVEN ELEMENT
'LONG
FLATTEN ENDS OF
TUBING AND DRILL
24 -8
RADIO
THE
Antennas
-F
heavy -duty soldering iron. The
Figure 12
SIMPLE 3- ELEMENT BEAM FOR
1'/ METERS
2
AND
driven element is next attached to the center
of the wood boom, mounted atop a small insulating plate made of bakelite, micarta or
some other non -conducting material. It is held
in place in the same manner as the parasitic
elements. The two free ends of the folded dipole are hammered flat and drilled for a 6 -32
bolt. These bolts pass through both the insulating block and the boom, and hold the free
tips of the element in place.
A length of 75 -ohm Twin -Lead TV -type line
should be used with this beam antenna. It is
connected to each of the free ends of the folded
dipole. If the.antenna is mounted in the vertical plane, the 75-ohm line should be brought
away from the antenna for a distance of four
to six feet before it drops down the tower to
lessen interaction between the antenna elements and the feed line. The complete antenna
is light enough to be turned by a TV rotator.
A simple Yagi antenna of this type will provide a gain of 7 db over the entire 2 -meter or
IS-meter band, and is highly recommended as
an "easy -to- build" beam for the novice or
beginner.
An 8- Element
"Tippoble" Array
Figures 13 and 14 illustrate an 8- element rotary
array for use on the 144 Mc. amateur band. This
array is "tippable" to obtain either horizontal
for 144 Mc.
or vertical polarization. It is necessary that
the transmitting and receiving station use the
same polarization for the ground -wave signal
propagation which is characteristic of this fre-
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HANDBOOK
VHF
Rhombic
483
TOP VIEW
0' TILT ANGLE
4).
6A
131.
Figure
10
RHOMBIC ANTENNA
The optimum
tilt
NON -INDUCTIVE
h
SIDE LENGTH, S
V -H -F
RI, R22390 OHMS EACH
1OA
Figure
DESIGN
CHART
angle (see figure
V -H -F
11
RHOMBIC ANTENNA
CONSTRUCTION
for
zero-angle" radiation depends upon the
length of the sides.
11)
10 to 16 db gain with a simpler construction
than does a phased dipole array, and has the
further advantage of being useful over a wide
frequency range.
Except at the upper end of the v -h -f range
a rhombic array having a worthwhile gain is
too large to be rotated. However, in locations
75 to 150 miles from a large metropolitan area
a rhombic array is ideally suited for working
into the city on extended (horizontally polarized) ground-wave while at the same time making an ideal antenna for TV reception.
The useful frequency range of a v -h -f rhombic array is about 2 to I, or about plus 40% and
minus 30% from the design frequency. This
coverage is somewhat less than that of a high frequency rhombic used for sky -wave communication. For ground -wave transmission or reception the only effective vertical angle is that
of the horizon, and a frequency range greater
than 2 to I cannot be covered with a rhombic
array without an excessive change in the vertical angle of maximum radiation or response.
The dimensions of a v -h -f rhombic array are
determined from the design frequency and figure 10, which shows the proper tilt angle (see
figure 11) for a given leg length. The gain of
a rhombic array increases with leg length.
There is not much point in constructing a v -h -f
rhombic array with legs shorter than about 4
wavelengths, and the beam width begins to become excessively sharp for leg lengths greater
than about 8 wavelengths. A leg length of 6
wavelengths is a good compromise between
beam width and gain.
The tilt angle given in figure 10 is based
upon a wave angle of zero degrees. For leg
lengths of 4 wavelengths or longer, it will be
necessary to elongate the array a few per cent
(pulling in the sides slightly) if the horizon
elevation exceeds about 3 degrees.
Table I gives dimensions for two dual purpose rhombic arrays. One covers the 6-meter
amateur band and the "low" television band.
The other covers the 2 -meter amateur band,
the "high" television band, and the 1%4-meter
amateur band. The gain is approximately 12
db over a matched half wave dipole and the
beam width is about 6 degrees.
The Feed Line
The recommended feed line
is an open -wire line having a
surge impedance between 450 and 600 ohms.
With such a line the VSWR will be less than
2 to 1. A line with two -inch spacing is suitable for frequencies below 100 Mc., but one inch spacing (such as used in the Gonset Line
for TV installations) is recommended for higher frequencies.
The Termination
If the array is to be used only
for reception, a suitable termination consists of two 390 -ohm carbon re-
6
METERS
AND LOW
TV
S
166'
L
(length)
W
-6
METERS
32'
10"
67' 4"
(Width)
METERS, NIGH
BAND TV, AND
2
11Q
90.
(side)
S
BAND
59' 4"
23' 11"
warelenths at design frequency
Tilt ongle 6B0
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TABLE I.
482
V -H -F
and
U -H
-F
THE
Antennas
RADIO
A
450 -ONM TV LINE
pA UHF
HORN ANTENNA
ANGLE BETWEEN
SIDES OF MORN '"606
D
OB
400
VHF HORIZONTALLY POLARIZED HORN
Figure
8
TYPES OF HORN ANTENNAS
The "two sided horn" of Figure BB may be
fed by means of on open -wire transmission
line.
i
ZA-A GAIN (DB)
430-OHM LINE
3
A
20
9
2a
390
1S
TWO SIDES MADE
OF WIRE MESH
Figure 9
THE 60° HORN ANTENNA FOR USE ON
FREQUENCIES ABOVE 144 MC.
TWO
Copper screen may also be used for the re-
flecting planes.
The values of spacing given in the corner reflector chart have been chosen such that the
center impedance of the driven element would
be approximately 70 ohms. This means that
the element may be fed directly with 70 -ohm
coaxial line, or a quarter-wave matching transformer such as a "Q" section may be used to
provide an impedance match between the center- impedance of the element and a 460 -ohm
line constructed of no. 12 wire spaced 2 inches.
In many v -h -f antenna systems, waveguide
transmission lines are terminated by pyramidal
horn antennas. These horn antennas (figure
8A) will transmit and receive either horizontally or vertically polarized waves. The use of
waveguides at 144 Mc. and 235 Mc., however,
is out of the question because of the relatively
large dimensions needed for a waveguide operating at these low frequencies.
A modified type of horn antenna may still be
used on these frequencies, since only one particular plane of polarization is of interest to
the amateur. In this case, the horn antenna
can be simplified to two triangular sides of
the pyramidal horn. When these two sides are
insulated from each other, direct excitation at
the apex of the horn by a two-wire transmission
line is possible.
In a normal pyramidal horn, all four triangular sides are covered with conducting material,
but when horizontal polarization alone is of
interest (as in amateur work) only the vertical
areas of the horn need be used. If vertical polarization is required, only the horizontal areas
of the horn are employed. In either case, the
system is unidirectional, away from the apex
of the horn. A typical horn of this type is shown
in figure 8B. The two metallic sides of the
horn are insulated from each other, and the
sides of the horn are made of small mesh
"chicken wire" or copper window screening.
A pyramidal horn is essentially a high -pass
device whose low frequency cut-off is reached
when a side of the horn is % wavelength. It
will work up to infinitely high frequencies,
the gain of the horn increasing by 6 db every
time the operating frequency is doubled. The
power gain of such a horn compared to a 1/2
wave dipole at frequencies higher than cutoff is:
8.4
A2
Power gain (db)
A2
where A is the frontal area of the mouth of the
horn. For the 60 degree horn shown in figure
8B the formula simplifies to:
Power gain (db) = 8.4 D2, when
pressed in terms of wavelength
D
is ex-
When D is equal to one wavelength, the power gain of the horn is approximately 9 db. The
gain and feed point impedance of the 60 degree horn are shown in figure 9. A 450 ohm
open wire TV -type line may be used to feed
the horn.
24 -7
VHF Horizontal
Rhombic Antenna
For v -h -f transmission and reception in a
fixed direction, a horizontal rhombic permits
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HANDBOOK
Helical
Antenna
Beam
22 in.
D
.16%2
S
in.
53 in.
G
Tubing o.d
1
in.
The D and S dimensions are to the center of
the tubing. These dimensions must be held
rather closely, since the range from 144 through
225 Mc. represents just about the practical
limit of coverage of this type of antenna system.
DRIVEN DIPOLE
SUPPORTING
ME
High -Band
TV Coverage
H
Note that an array constructed
with the above dimensions will
give unusually good high -band
TV reception in addition to covering the 144 Mc. and 220 -etc. amateur bands and the taxi
and police services.
On the 144 -Mc. band the beam width is approximately 60 degrees to the half -power
points, while the power gain is approximately
11 db over a non -directional circularly polarized antenna. For high -band TV coverage the
gain will be 12 to 14 db, with a beam width
of about 50 degrees, and on the 220 -Mc. amateur band the beam width will be about 40 degrees with a power gain of approximately 15 db.
The antenna system will receive vertically
polarized or horizontally polarized signals
with equal gain over its entire frequency range.
Conversely, it will transmit signals over the
same range, which then can be received with
equal strength on either horizontally polarized
vertically polarized receiving antennas.
The standing -wave ratio will be very low over
the complete frequency range if RG -63/U coaxial feed line is used.
or
24 -6
481
The Corner -Reflector
and Horn -Type Antennas
The corner -reflector antenna is a good directional radiator for the v -h -f and u -h -f region.
The antenna may be used with the radiating
element vertical, in which case the directivity
is in the horizontal or azimuth plane, or the
system may be used with the driven element
Figure 7
CONSTRUCTION OF THE "CORNER
REFLECTOR" ANTENNA
Such an antenna is capable of giving high
gain with a minimum of complexity in the
radiating system. It may be used either with
horizontal or vertical polarization. Design
data for the antenna is given in the Corner-
Reflector Design Table.
horizontal in which case the radiation is horizontally polarized and most of the directivity
is in the vertical plane. With the antenna used
as a horizontally polarized radiating system
the array is a very good low -angle beam array
although the nose of the horizontal pattern is
still quite sharp. When the radiator is oriented
vertically the corner reflector operates very
satisfactorily as a direction -finding antenna.
Design data for the corner -reflector antenna
is given in figure 7 and in the chart Cosner Re /lector Design Data. The planes which make
up the reflecting corner may be made of solid
sheets of copper or aluminum for the u -h -f
bands, although spaced wires with the ends
soldered together at top and bottom may be
used as the reflector on the lower frequencies.
CORNER- REFLECTOR DESIGN DATA
Corner
Angle
90
60
60
60
60
Freq.
Band, Mc.
R
110"
110"
38"
24.5"
13"
SO
50
144
220
420
NOTE:
H
S
82"
115"
140"
140"
40"
48"
25"
14"
30"
18"
Refer to figure
7
A
200"
230"
100"
72"
36"
Feed
L
G
Imped.
Approx.
Gain, db
230"
230"
100"
72"
18"
18"
72
70
70
70
70
10
12
12
12
12
36"
"
3"
5
for construction of corner- reflector an
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480
T
and
V -H -F
U -H
TRANSMIT
RECEIVE
,,,,-ROUND OR SQUARE
GROUND SCREEN
L
/\/\/\/\/\/\
G
THE
Antennas
-F
RADIO
used at a single frequency or over a narrow
band of frequencies, such as an amateur band.
At the design frequency the beam width is
about 50 degrees and the power gain about 12
db,referred to a non -directional circularly polarized antenna.
For the frequency range
100 to 500 Mc. a suitable
ground screen can be made from "chicken
wire" poultry netting of -inch mesh, fastened
to a round or square frame of either metal or
wood. The netting should be of the type that
is galvanized after weaving. A small, sheet
metal ground plate of diameter equal to approximately D/2 should be centered on the
screen and soldered to it. Tin, galvanized
iron, or sheet copper. is suitable. The outer
conductor of the RG -63/U (125 ohm) coax is
connected to this plate, and the inner conductor contacts the helix through a hole in the
center of the plate. The end of the coax should
be taped with Scotch electrical tape to keep
water out.
The Ground Screen
COAX FEED POINT (RG -63/U)
AT CENTER OF
GROUND SCREEN
t
D
=+
5=
á
G
=oer.
L.
i.
1
A
APPROX O.t1A
CONDUCTOR DIA
%= WAVELENGTH
IN FREE SPACE
Figure
6
THE "HELICAL BEAM" ANTENNA
This type of directional antenna system
gives excellent performance over o frequency
range of 1.7 to 1.8 to 1. Its dimensions are
such that it ordinarily is not practicable,
however, for use as a rotatable array on frequencies below about 100 Mc. The center
conductor of the feed line should pass
through the ground screen for connection to
the feed point. The outer conductor of the
coaxial line should be grounded to the
ground screen.
the time of writing, there has been no standardization of the "twist" for general amateur
work.
Perhaps the simplest antenna configuration
for a directional beam antenna having circular
polarization is the helical beam popularized
by Dr. John Kraus, W8JK. The antenna consists simply of a helix working against a
ground plane and fed with coaxial line. In the
u -h -f and the upper v -h -f range the physical
dimensions are sufficiently small to permit
construction of a rotatable structure without
much
difficulty.
the dimensions are optimized, the
characteristics of the helical beam antenna
are such as to qualify it as a broad band antenna. An optimized helical beam shows little
variation in the pattern of the main lobe and
a fairly uniform feed point impedance averagWhen
ing approximately 125 ohms over a frequency
range of as much as 1.7 to 1. The direction of
"electrical twist" (right or left handed) depends upon the direction in which the helix is
wound.
A six -turn helical beam is shown schematically in figure 6. The dimensions shown will
give good performance over a frequency range
of plus or minus 20 per cent of the design frequency. This means that the dimensions are
not especially critical when the array is to be
It should
be noted that the beam
proper consists of six full turns.
The start of the helix is spaced a distance of
S/2 from the ground screen, and the conductor
goes directly from the center of the ground
screen to the start of the helix.
Aluminum tubing in the "SO" (soft) grade
is suitable for the helix. Alternatively, lengths
of the relatively soft aluminum electrical conduit may be used. In the v -h -f range it will be
necessary to support the helix on either two
or four wooden longerons in order to achieve
sufficient strength. The longerons should be
of as small cross section as will provide sufficient rigidity, and should be given several
coats of varnish. The ground plane butts
against the longerons and the whole assembly
is supported from the balance point if it is to
be rotated.
Aluminum tubing in the larger diameters ordinarily is not readily available in lengths greater than 12 feet. In this case several lengths
can be spliced by means of short telescoping
sections and sheet metal screws.
The tubing is close wound on a drum and
then spaced to give the specified pitch. Note
that the length of one comp e t e turn when
spaced is somewhat greater than the circumference of a circle having the diameter D.
The Helix
1
Broad -Band
144 to 225 Mc.
Helical Beam
A highly useful v -h -f helical
beam which will receive sig-
nals with good gain over the
complete frequency range from
144 through 225 Mc. may be constructed by
using the following dimensions (180 Mc. design center):
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HANDBOOK
Discone
0.1
479
Antenna
D
400i
300
200
160
160
140
120'
110
MO'
90
so
50n. COAX
(PIG-4/U, ETC.)
To
60
Figure SA
THE ''DISCONE " BROAD -BAND
RADIATOR
This antenna system radiates a vertically
polarized wave over a very wide frequency
range. The "disc" may be made of solid
met al sheet, a group of radials, or wire
screen; the "cone" may best be constructed
by forming a sheet of thin aluminum. A single antenna may be used for operation on the
50,
144, and 220 Mc.
amdteur bands. The
dimension D is determined by the lowest frequency to be employed, and is given in the
50
O.!
t 0
15
DIN
DESIGN
2
2.5
3
4
6
FEET
Figure 5B
CHART FOR THE "DISCONE"
ANT ENN A
of the skirt directly to an effective ground
plane such as the top of an automobile.
chart of figure 58.
24 -5
Helical Beam
Antennas
VSXRof less than 1.5 will be obtained throughout the operating range of the antenna.
The Discone antenna may be considered
as a cross between an electromagnetic horn
and an inverted ground plane unipole antenna.
It looks to the feed line like a properly terminated high -pass filter.
Construction Details
The top disk and the
conical skirt may be
fabricated either from sheet metal, screen (such
as "hardware cloth "), or 12 or more "spine"
radials. If screen is used a supporting framework of rod or tubing will be necessary for
mechanical strength except at the higher frequencies.. If spines are used, they should be
terminated on a
ring for mechanical
strength except at the higher frequencies.
The top disk is supported by means of three
insulating pillars fastened to the skirt. Either
polystyrene or low -loss ceramic is suitable for
the purpose. The apex of the conical skirt is
grounded to the supporting mast and to the
outer conductor of the coaxial line. The line
is run down through the supporting mast. An
alternative arrangement, one suitable for certain mobile applications, is to fasten the base
stiff
Most v -h -f and
u -h -f antennas are either verpolarized or horizontally polarized
(plane polarization). However, circularly polarized antennas have interesting characteristics which may be useful for certain applications. The installation of such an antenna can
effectivèly solve the problem of horizontal vs.
vertical polarization.
A circularly polarized wave has its energy
divided equally between a vertically polarized
component and a horizontally polarized component, the two being 90 degrees out of phase.
The circularly polarized wave may be either
"left handed" or "right handed," depending
upon whether the vertically polarized component leads or lags the horizontal component.
A circularly polarized antenna will respond
to any plane polarized wave whether horizontally polarized, vertically polarized, or diagonally polarized. Also, a circular polarized
wave can be received on a plane polarized antenna, regardless of the polarization of the
latter. When using circularly polarized antennas at both ends of the circuit, however, both
must be left handed or both must be right
handed. This offers some interesting possibilities with regard to reduction of QRM. At
tically
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478
V -H -F
and
U
-H -F
Antennas
THE
f
1
36"
TYP.
RADIO
jALUMINUM TUBING
.
2X2 191 IR'
TYP.
220 MC.
w 19"
T
I
S.I.
=
ak-
E/r,
..UU/MUMCM7SSBAB TIGHTENS IT UP.
300-OHM
FEEDLIN
TOP APEX CONNECTS TO
INNER CONNECTOR
LOWER APEX CONNECTS
TO OUTER CONDUCTOR
APICES FORMED
~ -OF
SHEET METAL
300-OHM
TUBULAR
TWIN LEAD
20'
300-OHM
FEEDLINE
RD-B /U CABLE
Figure
3
THE DOUBLE SKELETON CONE
ANTENNA
A skeleton cone has been substituted for the
single element radiator of figure 2C. This
greatly increases the bandwidth. If at least
10 elements are used for each skeleton cone
and the angle of revolution and element
length are optimized, a low SWR con be obtained over o frequency range of at least two
octaves. To obtain this order of bandwidth,
the element length L should be approximately 0.2 wavelength at the lower frequency end
of the band, and the angle of revolution optimized for the lowest maximum VSWR within
the frequency range to be covered. A greater
improvement in the impedance -frequency
characteristic can be achieved by adding
elements than by increasing the diameter of
the elements. With only 3 elements per
"cone'. and a much smaller angle of revolution a low SWR can be obtained over a frequency range of approximately 1.3 to 1.0
when the element lengths are optimized.
Figure 4
NONDIRECTIONAL ARRAYS FOR
use.
acing frequency. The antenna then will perform well over a frequency range of at least
At certain frequencies within this
range the vertical pattern will tend to "lift"
slightly, causing a slight reduction in gain at
zero angular elevation, but the reduction is
very slight.
Below the frequency at which the slant
height of the conical skirt is equal to a free space quarter wavelength the standing-wave
ratio starts to climb, and below a frequency
approximately 20 per cent lower than this the
standing -wave ratio climbs very rapidly. This
is termed the cut off frequency of the antenna.
By making the slant height approximately equal
to a free -space quarter wavelength at the lowest frequency employed (refer to chart), a
8 to 1.
work over several octaves, the gain varying
only slightly over a very wide frequency range.
Commercial versions of the Discone antenna for various applications are manufactured
by the Federal Telephone and Radio Corporation. A Discone type antenna for amateur work
can be fabricated from inexpensive materials
with ordinary hand tools.
A Discone antenna suitable for multi -band
amateur work in the v- h /u-h -f range is shown
schematically in figure 5A. The distance D
should be made approximately equal to a free space quarter wavelength at the lowest oper-
144 MC.
AND 235 MC.
On right is shown two band installation. The
whole system may easily be dissembled and
carried on a ski -rock atop a car for portable
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Vertically Polarized Arrays
HANDBOOK
CLOSED
I
r
~OPEN
Figure 2
THREE VERTICALLY -POLARIZED
LOW -ANGLE RADIATORS
Shown at (A) is the "sleeve" or "'hypodermic" type of radiator. At (©) is shown the
ground-plane vertical, and (C) shows a modification of this antenna system which increases the feed-point impedance to a value
such that the system may be fed directly
from o coaxial line with no standing waves
on the feed line.
matching transformer, and a good match is
obtained.
In actual practice the antenna would consist of a quarter-wave rod, mounted by means
of insulators atop a pole or pipe mast. Elaborate insulation is not required, as the voltage
at the lower end of the quarter -wave radiator
is very low. Self- supporting rods from 0.25 to
0.28 wavelength would be extended out, as in
the illustration, and connected together. As
the point of connection is effectively at ground
potential, no insulation is required; the horizontal rods may be bolted directly to the supporting pole or mast, even if of metal. The coaxial line should be of the low loss type especially designed for v -h -f use. The outside
connects to the junction of the radials, and
the inside to the bottom end of the vertical
radiator. An antenna of this type is moderately
simple to construct and will give a good account of itself when fed at the lower end of the
radiator directly by the 52 -ohm RG -8 /U coaxial cable. Theoretically the standing -wave
ratio will be approximately 1.5-to -1 but in
practice this moderate s -w -r produces no
deleterious effects, even on coaxial cable.
The modification shown in figure 2C permits
matching to a standard 50- or 70 -ohm flexible
coaxial cable without a linear transformer. If
the lower rods hug the line and supporting mast
477
rather closely, the feed -point impedance is
about 70 ohms. If they are bent out to form an
angle of about 30° with the support pipe the
impedance is about 50 ohms.
The number of radial legs used in a ground plane antenna of either type has an important
effect on the feed -point impedance and upon
the radiation characteristics of the antenna
system. Experiment has shown that three radials is the minimum number that should be
used, and that increasing the number of radials above six adds substantially nothing to the
effectiveness of the antenna and has no effect
on the feed -point impedance. Experiment has
shown, however, that the radials should be
slightly longer than one -quarter wave for best
results. A length of 0.28 wavelength has been
shown to be the optimum value. This means
that the radials for a 50 -Mc. ground -plane vertical antenna should be 65" in length.
The bandwidth of the antenna of figure 2C can be increased considerably by substituting several space -tapered rods for the
single radiating element, so that the "radiator" and skirt are similar. If a sufficient number of rods are used in the skeleton cones and
the angle of revolution is optimized for the
particular type of feed line used, this antenna
exhibits a very low SWR over a 2 to 1 frequency range. Such an arrangement is illustrated
schematically in figure 3.
Double Skeleton
Cone Antenna
Nondirectional
Vertical Array
Half-wave elements may be
stacked in the vertical plane
to provide a non -directional
pattern with good horizontal gain. An array
made up of four half -wave vertical elements
is shown in figure 4A. This antenna provides
a circular pattern with a gain of about 4.5 db
over a vertical dipole. It may be fed with
300 -ohm TV -type line. The feedline should be
conducted in such a way that the vertical portion of the line is at least one-half wavelength
away from the vertical antenna elements. A
suitable mechanical assembly is shown in figure 4B for the 144 -Mc. and 235 -Mc. amateur
bands.
A
24 -4
The Discone Antenna
The Discone antenna is a vertically polarized omnidirectional radiator which has very
broad band characteristics and permits a simple, rugged structure. This antenna presents a
substantially uniform feed -point impedance,
suitable for direct connection of a coaxial
line, over a range of several octaves. Alsg,
the vertical pattern is suitable for ground -wave
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476
V -H -F
and
U
-H -F
RADIO
THE
Antennas
1
1
2=70n
VECTOR SUM OF
2 PATTERNS
COAXIAL LINE
TO TRANSMITTER
LOW Z
TRANSMISSION LINE
TO XMTR
O
Figure
HORIZONTALLY
©
1
THREE
NONDIRECTIONAL,
radiation at the very low elevation angles are
not recommended for v -h -f and u -h -f work. It is
for this reason that the horizontal dipole and
horizontally- disposed colinear arrays are generally unsuitable for work on these frequencies. Arrays using broadside or end-fire elements do concentrate radiation at low elevation angles and are recommended for v -h-f
work. Arrays such as the lazy -H, Sterba curtain, flat -top beam, and arrays with parasitically excited elements are recommended for
this work. Dimensions for the first three types
of arrays may be determined from the data
given in the previous chapter, and reference
may be made to the Table of Wavelengths given
in this chapter.
Arrays using vertically- stacked horizontal
dipoles, such as are used by commercial television and FM stations, are capable of giving
high gain without a sharp horizontal radiation
pattern. If sets of crossed dipoles, as shown
in figure 1A, are fed 90° out of phase the resulting system is called a turnstile antenna.
The 90° phase difference between sets of dipoles may be obtained by feeding one set of
dipoles with a feed line which is one -quarter
wave longer than the feed line to the other
set of dipoles. The field strength broadside to
one of the dipoles is equal to the field from
that dipole alone. The field strength at a point
at any other angle is equal to the vector sum
of the fields from the two dipoles at that angle. A nearly circular horizontal pattern is
produced by this antenna.
A second antenna producing a uniform, horizontally polarized pattern is shown in figure
1B. This antenna employs three dipoles bent
to form a circle. All dipoles are excited in
phase, and are center fed. A bazooka is included in the system to prevent unbalance in
the coaxial feed system.
POLARIZED
ANTENNAS
A third nondirectional antenna is shown in
figure IC. This simple antenna is made of two
half-wave elements, of which the end quarter wavelength of each is bent back 90 degrees.
The pattern from this antenna is very much
like that of the turnstile antenna. The field
from the two quarter -wave sections that are
bent back are additive because they are 180
degrees out of phase and are a half wavelength apart. The advantage of this antenna is
the simplicity of its feed system and construction.
24 -3
Simple Vertical -Polarized
Antennas
For general coverage with a single antenna,
single vertical radiator is commonly employed. A two -wire open transmission line is
not suitable for use with this type antenna,
and coaxial polyethylene feed line such as
RG-8 /U is to be recommended. Three practical
methods of feeding the radiator with concentric line, with a minimum of current induced
in the outside of the line, are shown in figure
2. Antenna (A) is known as the sleeve antenna, the lower half of the radiator being a large
piece of pipe up through which the concentric
feed line is run. At (B) is shown the ground plane vertical, and at (C) a modification of
this latter antenna.
resistance of the ground The radiation
plane vertical is approximately 30 ohms, which
is not a standard impedance for coaxial line.
To obtain a good match, the first quarter wavelength of feeder may be of 52 ohms surge impedance, and the remainder of the line of approximately 75 ohms impedance. Thus, the
first quarter -wave section of line is used as a
a
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HANDBOOK
Antenna Polarization
There is no point in using
copper tubing for an antenna
on the medium frequencies.
The reason is that considerable tubing would
be required, and the cross section still would
not be a sufficiently large fraction of a wavelength to improve the antenna bandwidth characteristics. At very high and ultra high frequencies, however, the radiator length is so
short that the expense of large diameter conductor is relatively small, even though copper
pipe of 1 inch cross section is used. With such
conductors, the antenna will tune much more
broadly, and often a broad resonance characteristic is desirable. This is particularly true
when an antenna or array is to be used over
an entire amateur band.
It should be kept in mind that with such
large cross section radiators, the resonant
length of the radiator will be somewhat shorter,
being only slightly greater than 0.90 of a half
wavelength for a dipole when heavy copper
pipe is used above 100 Mc.
Radiator Cross
Section
The matter of insulation is of
prime importance at very high frequencies. Many insulators that have very low
losses as high as 30 Mc. show up rather poorly at frequencies above 100 Mc. Even the low
loss ceramics are none too good where the r -f
voltage is high. One of the best and most practical insulators for use at this frequency is
polystyrene. It has one disadvantage, however,
in that it is subject to fracture and to deformation in the presence of heat.
It is common practice to design v -h -f and
u -h -f antenna systems so that the various radiators are supported only at points of relatively
low voltage; the best insulation, obviously, is
air. The voltages on properly operated untuned
feed lines are not high, and the question of
insulation is not quite so important, though insulation still should be of good grade.
Insulation
475
TABLE OF WAVELENGTHS
Fra.
quency
in Mc.
t/4
Wove
Free
Space
50.0
50.5
51.0
51.5
52.0
52.5
53.0
54.0
58.5
57.9
57.4
56.8
56.3
55.7
54.7
144
145
146
147
148
235
236
237
238
239
240
Wave
Anrenna
55.5
55.0
54.4
53.9
53.4
1/2
Wave
Free
Space
1/2
Wane
An-
renna
51 .4
109.5
111.0
109.9
108.8
107.8
106.7
105.7
104.7
102.8
20.5
20.4
20.2
20.0
19.9
19.2
18.9
18.8
18.6
41.0
40.8
40.4
40.0
39.9
38.5
38.3
38.0
37.6
37.2
12.6
12.5
12.5
12.4
12.4
12.3
11.8
11.8
11.7
11.7
11.6
11.6
25.2
25.1
25.0
24.9
24.8
24.6
23.6
23.5
23.5
23.4
23.3
23.2
14.1
13.25
S9.1
420
425
430
1/4
7.05
6.95
6.88
5 2. 8
52 4
19.1
6.63
6.55
6.48
118.1
116.9
115.9
114.7
113.5
112.5
1
1
1
.5
13.9
13.8
13.1
12.95
All dimensions ore in inches. Lengths hove in
most cases been rounded off to three significant
figures. "1/2 -Wave Free -Space' column shown
above should be used with Lecher wires for frequency measurement.
.
Antenna
Commercial broadcasting in the
U.S.A. for both FM and television in the v -h -f range has
been standarized on horizontal polarization.
One of the main reasons for this standardization is the fact that ignition interference is
reduced through the use of a horizontally polarized receiving antenna. Amateur practice,
however, is divided between horizontal and
vertical polarization in the v -h-f and u -h -f
range. Mobile stations are invariably vertical cally polarized due to the physical limitations
imposed by the automobile antenna installation. Most of the stations doing intermittent
or occasional work on these frequencies use a
simple ground-plane vertical antenna for both
transmission and reception. However, those
Polarization
stations doing serious work and striving for
maximum -range contacts on the 50 -Mc. and
144 -Mc. bands almost invariably use horizon-
tal polarization.
Experience has shown that there is a great
attenuation in signal strength when using
crossed polarization (transmitting antenna
with one polarization and receiving antenna
with the other) for all normal ground -wave contacts on these bands. When contacts are being made through sporadic -E reflection, however, the use of crossed polarization seems to
make no discernible difference in signal
strength. So the operator of a station doing
v -h -f work (particularly on the 50 -Mc. band)
is faced with a problem: If contacts are to be
made with all stations doing work on the same
band, provision must be made for operation on
both horizontal and vertical polarization. This
problem has been solved in many cases through
the construction of an antenna array that may
be revolved in the plane of polarization in addition to being capable of .rotation in the azimuth plane.
An alternate solution to the problem which
involves less mechanical construction is simply to install a good ground -plane vertical antenna for all vertically- polarized work, and
then to use a multi -element horizontally- polarized array for dx work.
24 -2
Simple Horizontally -
Polarized Antennas
Antenna systems which do not concentrate
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474
V -H -F
and
U -X -F
THE
Antennas
RADIO
that both are directed at the station being received. Many instances have been reported
where a frequency band sounded completely
dead with a simple dipole receiving antenna
but when the receiver was switched to a three element or larger array a considerable amount
of activity from 80 to 160 miles distant was
heard.
of about 50 feet or less. For longer runs, either
the u -h -f or v -h -f TV open -wire lines may be
used with good overall efficiency. The v -h -f
line is satisfactory for use on the amateur
420 -Mc. band.
Angle of
connection, however, is the antenna changeover relay. Reflections at the antenna changeover relay become of increasing importance
as the frequency of transmission is increased.
When coaxial cable is used as the antenna
transmission line, satisfactory coaxial antenna changeover relays with low reflection can
be used. One type manufactured by Advance
Electric & Relay Co., Los Angeles 26, Calif.,
will give a satisfactorily low value of reflection.
On the 235-Mc. and 420 -Mc. amateur bands,
the size of the antenna array becomes quite
small, and it is practical to mount two identical antennas side by side. One of these antennas is used for the transmitter, and the
other antenna for the receiver. Separate transmission lines are used, and the antenna relay
may be eliminated.
The useful portion of the signal
in the v -h -f and u -h -f range for
short or medium distance communication is that which is radiated at a very low
angle with respect to the surface of the earth;
essentially it is that signal which is radiated
parallel to the surface of the earth. A vertical
antenna transmits a portion of its radiation at
a very low angle and is effective for this reason; its radiation is not necessarily effective
simply because it is vertically polarized. A
simple horizontal dipole radiates very little
low-angle energy and hence is not a satisfactory v -h -f or u -h -f radiator. Directive arrays
which concentrate a major portion of the radiated signal at a low radiation angle will prove
to be effective radiators whether their signal
is horizontally or vertically polarized.
In all cases, the radiating system for v -h -f
and u -h -f work should be as high and in the
clear as possible. Increasing the height of the
antenna system will produce a very marked
improvement in the number and strength of the
signals heard, regardless of the actual type
of antenna used.
Radiation
Transmission lines to v -h -f and
u -h -f antenna systems may be
either of the parallel- conductor
or coaxial conductor type. Coaxial line is recommended for short runs and closely spaced
open -wire line for longer runs. Wave guides
may be used under certain conditions for frequencies greater than perhaps 1500 Mc. but
their dimensions become excessively great for
frequencies much below this value. Non- resonant transmission lines will be found to be considerably more efficient on these frequencies
than those of the resonant type. It is wise to
to use the very minimum length of transmission
line possible since transmission line losses
at frequencies above about 100 Mc. mount very
rapidly.
Open sines should preferably be spaced
closer than is common for longer wavelengths,
as 6 inches is an appreciable fraction of a
wavelength at 2 meters. Radiation from the
line will be greatly reduced if 1 -inch or 11/4inch spacing is used, rather than the more common 6-inch spacing.
Ordinary TV-type 300 -ohm ribbon may be
used on the 2 -meter band for feeder lengths
Transmission
Lines
It is recommended that the same
antenna be used for transmitting
and receiving in the v -h -f and
u -h -f range. An ever- present problem in this
Antenna
Changeover
vertical radiator for
general coverage u -h -f
use should be made
either 1/4 or % wavelength
long. Longer vertical antennas do not have
their maximum radiation at right angles to the
line of the radiator (unless co- phased), and,
therefore, are not practicable for use where
greatest possible radiation parallel to the
earth is desired.
Unfortunately, a feed system which is not
perfectly balanced and does some radiating,
not only robs the antenna itself of that much
power, but distorts the radiation pattern of the
antenna. As a result, the pattern of a vertical
radiator may be so altered that the radiation
is bent upwards slightly, and the amount of
power leaving the an t e n n a parallel to the
earth is greatly reduced. A vertical half -wave
radiator fed at the bottom by a quarter -wave
stub is a good example of this; the slight
radiation from the matching section decreases
the power radiated parallel to the earth by
nearly 10 db.
The only cure is a feed system which does
not disturb the radiation pattern of the antenna
itself. This means that if a 2 -wire line is used,
the current and voltages must be exactly the
same (though 180° out of phase) at any point
on the feed line. It means that if a concentric
feed line is used, there should be no current
flowing on the outside of the outer conductor.
Effect of Feed
System on Radiation
Angle
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A
CHAPTER TWENTY -FOUR
V-li-F
and
U-li-F
The very- high -frequency or v -h -f frequency
range is defined as that range falling between
30 and 300 Mc. The ultra- high -frequency or
u -h -f range is defined as falling between 300
and 3000 Mc. This chapter will be devoted to
the design and construction of antenna systems for operation on the amateur 50 -Mc., 144 Mc., 235 -Mc., and 420 -Mc. bands. Although the
basic principles of antenna operation are the
same for all frequencies, the shorter physical
length of a wave in this frequency range and
the differing modes of signal propagation make
it possible and expedient to use antenna systems different in design from those used on
the range from 3 to 30 Mc.
24 -1
Antennas
station. Even a much simpler and smaller three or four -element parasitic array having a gain
of 7 to 10 db will produce a marked improvement in the received signal at the other station.
11 o w e v e r,
as all v -h -f and u -h -f workers
know, the most important contribution of a
high -gain antenna array is in reception. If a
remote station cannot be heard it obviously is
impossible to make contact. The limiting factor in v -h -f and u -h -f reception is in almost
every case the noise generated within the receiver itself. Atmospheric noise is almost nonexistent and ignition interference can almost
invariably be reduced to a satisfactory level
through the use of an effective noise limiter.
Even with a grounded -grid or neutralized triode
first stage in the receiver the noise contribution of the first tuned circuit in the receiver
will be relatively large. Hence it is desirable
to use an antenna system which will deliver
the greatest signal voltage to the first tuned
circuit for a given field strength at the receiving location.
Antenna Requirements
Any type of antenna system useable on the
lower frequencies may be used in the v -h -f and
u -h -f bands. In fact, simple non -directive half wave or quarter -wave vertical antennas are
very popular for general transmission and reception from all directions, especially for
short-range work. But for serious v -h -f or u -h -f
work the use of some sort of directional antenna array is a necessity. In the first place,
when the transmitter power is concentrated into a narrow beam the apparent transmitter power at the receiving station is increased many
times. A "billboard" array or a Sterba curtain
having a gain of 16 db will make a 25 -watt
transmitter sound like a kilowatt at the other
Since the field intensity being produced at
the receiving location by a remote transmitting
station may be assumed to be constant, the receiving antenna which intercepts the greatest
amount of wave front, assuming that the polarization and directivity of the receiving antenna
is proper, will be the antenna which gives the
best received signal -to-noise ratio. An antenna
which has two square wavelengths effective
area will pick up twice as much signal power
as one which has one square wavelength area,
assuming the same general type of antenna and
473
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472
High Frequency Directive Antennas
Thus it is seen that, while maximum gain
occurs with two stacked dipoles at a spacing
of about 0.7 wavelength and the space directivity gain is approximately 5 db over one element under these conditions; the case of two
flat top or parasitic arrays stacked one above
the other is another story. Maximum gain will
occur at a greater spacing, and the gain over
one array will not appreciably exceed 3 db.
When two broadside curtains are placed one
ahead of the other in end -fire relationship, the
aggregate mutual impedance between the two
curtains is such that considerable spacing is
required in order to realize a gain approaching
3 db (the required spacing being a function of
the size of the curtains). While it is true that
a space directivity gain of approximately 4 db
can be obtained by placing one, half -wave dipole an eighth wavelength ahead of another
and feeding them 180 degrees out of phase, a
gain of less than 1 db is obtained when the
same procedure is applied to two large broadside curtains. To obtain a gain of approximately 3 db and retain a bidirectional pattern, a
spacing of many wavelengths is required between two large curtains placed one ahead of
the other.
A different situation exists, however, when
one driven curtain is placed ahead of an identical one and the two are phased so as to give
a unidirectional pattern. When a unidirectional
pattern is obtained, the gain over one curtain
will be approximately 3 db regardless of the
spacing. For instance, two large curtains
placed one a quarter wavelength ahead of the
other may have a space directivity gain of only
0.5 db over one curtain when the two are driven 180 degrees out of phase to give a bidirectional pattern (the type of pattern obtained
with a single curtain). However, if they are
driven in phase quadrature (and with equal currents) the gain is approximately 3 db.
The directivity gain of a composite array
also can be explained upon the basis of the
directivity patterns of the component arrays
alone, but it entails a rather complicated picture. It is sufficient for the purpose of this
discussion to generalize and simplify by saying that the greater the directivity of an end fire array, the farther an identical array must
be spaced from it in broadside relationship to
obtain optimum performance; and the greater
the directivity of a broadside array, the farther
an identical array must be spaced from it in
end -fire relationship to obtain optimum performance and retain the bidirectional charac-
teristic.
It is important to note that while a bidirectional end -fire pattern is obtained with two
driven dipoles when spaced anything under a
half wavelength, and while the proper phase
relationship is 180 degrees regardless of the
spacing for all spacings not exceeding one
half wavelength, the situation is different in
the case of two curtains placed in end -fire relationship to give a bidirectional pattern. For
maximum gain at zero wave angle, the curtains
should be spaced an odd multiple of one half
wavelength and driven so as to be 180 degrees
out of phase, or spaced an even multiple of
one half wavelength and driven in the same
phase. The optimum spacing and phase relationship will depend upon the directivity pattern of the individual curtains used alone, and
as previously noted the optimum spacing increases with the size and directivity of the
component arrays.
A concrete example of a combination broadside and end -fire array is two Lazy H arrays
spaced along the direction of maximum radiation by a distance of four wavelengths and fed
in phase. The space directivity gain of such
an arrangement is slightly less than 9 db. However, approximately the same gain can be obtained by juxtaposing the two arrays side by
side or one over the other in the same plane,
so that the two combine to produce, in effect,
one broadside curtain of twice the area. It is
obvious that in most cases it will be more expedient to increase the area of a broadside
array than to resort to a combination of end fire and broadside directivity. One exception,
of course, is where two curtains are fed in
phase quadrature to obtain a unidirectional
pattern and space directivity gain of approximately 3 db with a spacing between curtains
as small as one quarter wavelength. Another
exception is where very low angle radiation is
desired and the maximum pole height is strictly limited. The two aforementioned Lazy H
arrays when placed in end-fire relationship
will have a considerably lower radiation angle
than when placed side by side if the array elevation is low, and therefore may under some
conditions exhibit appreciably more practical
signal gain.
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HANDBOOK
Triplex
Beam
471
'
RO.[ "
Figure 24
THE TRIPLEX FLATTOP BEAM
ANTENNA FOR 10, 15 AND 20
METERS
S
u-
11
MAXIMUM
MAX. RADIATION
RADIATION
4.S DS
4.5
3000 LINE
OR
TO
TRANSMITTER
ANY LENGTH
DIMENSIONS
to one -quarter wave spacing may be used on
the fundamental for the one -section types and
also the two -section center-fed, but it is not
desirable to use more than 0.15 wavelength
spacing for the other types.
Although the center -fed type of flat -top generally is to be preferred because of its symmetry, the end -fed type often is convenient or
desirable. For example, when a flat -top beam
is used vertically, feeding from the lower end
is in most cases more convenient.
If a multisection flat -top array is end -fed
instead of center-fed, and tuned feeders are
used, stations off the ends of the array can be
worked by tying the feeders together and working the whole affair, feeders and all, as a long wire harmonic antenna. A single -pole double throw switch can be used for changing the
feeders and directivity.
The Triplex
The Triplex beam is
a modified
of the W8] K antenna
which uses folded dipoles for
the half wave elements of the array. The use
of folded dipoles results in higher radiation
resistance of the array, and a high overall system performance. Three wire dipoles are used
for the elements, and 300 -ohm Twin -Lead is
Beam
version
10M.
15M. 20M
MATERIAL
L
1'S
21'5' 32'2
iEL[tA[b 3'
S
5'0.
7'11.
II'
D
7'2'
10'7'
14'4" 3000M RIeeON
used for the two phasing sections. A recommended assembly for Triplex beams for 28 Mc.,
21 Mc., and 14 Mc. is shown in figure 24. The
gain of a Triplex beam is about 4.5 db over a
dipole.
23 -8
Combination End -Fire and
Broadside Arrays
Any of the end -fire arrays previously described may be stacked one above the other or
placed end to end (side by side) to give greater directivity gain while maintaining a bidirectional characteristic. However, it must be
kept in mind that to realize a worthwhile increase in directivity and gain while maintaining a bidirectional pattern the individual arrays must be spaced sufficiently to reduce the
mutual impedances to a negligible value.
When two flat top beams, for instance, are
placed one above the other or end to end, a
center spacing on the order of one wavelength
is required in order to achieve a worthwhile
increase in gain, or approximately 3 db.
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THE
High Frequency Directive Antennas
470
CENTER FED
RADIO
END FED
TO CENTER
FLAT TOP
or
I.-A-4
It
r
MATCHING STUB
M
L,
{-oi
1- SECTION
1- SECTION
,
L,
11
A
--{
1
-L2 -'"i
1-IX
2- SECTION
-L2
g
.---L3
01-
L2--
t
fol
La
12M
CONNCCF ATCr
L3I
l'L3
STUB
ERS
3-SECTION
r
S
M
L2--1
Ioi
I---La
loi
4-SECTION
g
r
S
L3-
L4
2M
S
M
,
I--- L3- 1-o- -- L3- 4pr.- L3-404-- L3-
k-- L3-4 1_La _J-4-
La ----1-61--1-3-.4
FIGURE 23
FLAT -TOP BEAM (8JK ARRAY) DESIGN DATA.
FREQUENCY
Spar.
'os
7.0-7.2 Mc. X/8
7.2 -7.3
14.0 -14.4
14.0 -14.4
14.0 -14.4
14.0 -14.4
28.0-29.0
28.0 -29.0
29.0 -30.0
29.0 -30.0
a/8
)/B
A
S
L.
L,
17'4' 34
17'0' 33'6'
8'8'
17'
10'5' 17'
13'11' 17'
a/4 17'4' 17'
.15)
5'2' 8'6'
.15X
.20X
a/4
.15X
X/4
8'8'
5'0'
8'4'
8'6'
8'3'
8'3'
60'
59'
30'
30'
30'
30'
L,
L.
52'8'
51'8'
44'
26'4
22'
43'1'
M
D
8'10' 4'
8'8' 4'
4'S' 2'
2S'3' 20'
5'4' 2'
22'10'
7'2' 2'
20'8'
8'10' 2'
15'
12'7' 10'
2'8' 1'6'
15'
10'4'
4'S' 1'6'
14'6' 12'2' 9'8'
2'7' 1'6'
14'6' 10'0'
4'4' 1'6'
(/)
A ('A)
A (1/4)
X
60'
59'
30'
29'
27'
25'
96'
94'
48'
47'
45'
43'
24'
22'
23'
21'
4'
4'
approx. approx. approx. approx.
26'
26'
13'
12'
10'
8'
7'
5'
7'
5'
15'
13'
15'
13'
2'
2'
3'
4'
1'
2'
1'
2'
Dimension chart for flat -top beam antennas. The meanings of the symbols are as fo lows:
L. L. and L,, the lengths of the sides of the flat-top sections as shown. L, is length
of the sides of single -section center -fed, L. single- section end -fed and 2- section center -fed, L, 4- section
center -fed and end -sections of 4- section end -fed, and L, middle sections of 4- section end -fed.
S, the spacing between the flat -top wires.
M, the wire length from the outside to the center of each cross -over.
D, the spacing lengthwise between sections.
A (1/4), the approximate length for a quarter-wave stub.
A (''s), the approximate length for a half -wave stub.
A (3/4), the approximate length for a three -quarter wave stub.
X, the approximate distance above the shorting wire of the stub for the connection of a 600 -ohm
line. This distance, as given in the table, is approximately correct only for 2- section flat-tops.
For single- section types it will be smaller and for 3- and 4- section types it will be larger.
The lengths given for a half -wave stub are applicable only to single -section center -fed flat-tops. To
be certain of sufficient stub length, it is advisable to make the stub a foot or so longer than shown in
the table, especially with the end -fed types. The lengths, A, are measured from the point where the
stub connects to the flat-top.
Both the center and end -fed types may be used horizontally. However, where a vertical antenna is
desired, the flat -tops can be turned on end. In this case, the end -fed types may be more convenient,
feeding from the lower end.
L
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HANDBOOK
Endlire Arrays
Normally the antenna tank will be located
as the transmitter, to facilitate adjustment when changing frequency. In
this case it is recommended that the link coupled tank be located across the room from the
transmitter if much power is used, in order to
minimize r -f feedback difficulties which might
occur as a result of the asymmetrical high impedance feed. If tuning of the antenna tank
from the transmitter position is desired, flexible shafting can be run from the antenna tank
condenser to a control knob at the transmitter.
The lower end of the driven element is quite
"hot" if much power is used, and the lead -in
insulator should be chosen with this in mind.
The ground connection need not have very low
resistance, as the current flowing in the
ground connection is comparatively small. A
stake or pipe driven a few feet in the ground
will suffice. However, the ground lead should
be of heavy wire and preferably the length
should 'not exceed about 10 feet at 7 Mc. or
about 20 feet at 4 Mc. in order to minimize
reactive effects due to its inductance. If it is
impossible to obtain this short a ground lead,
a piece of screen or metal sheet about four
feet square may be placed parallel to the earth
in a convenient location and used as an artificial ground. A fairly high C/L ratio ordinarily will be required in the antenna tank in order
to obtain adequate coupling and loading.
in the same room
23 -7
End -Fire
Directivity
By spacing two half-wave dipoles, or colinear arrays, at a distance of from 0.1 to 0.25
wavelength and driving the two 180° out of
phase, directivity is obtained through the two
wires at right angles to them. Hence, this type
of bidirectional array is called end fire. A better idea of end -fire directivity can be obtained
by referring to figure 10.
Remember that end-fire refers to the radiation with respect to the two wires in the array
rather than with respect to the array as a
whole.
The vertical directivity of an end -fire bidirectional array which is oriented horizontally can be increased by placing a similar end fire array a half wave below it, and excited in
the same phase. Such an array is a combination broadside and end -fire affair.
Flat -Top
very effective bidirectional
end -fire array is the Kraus or
8JK Hai-Top Beam. Essentially, this antenna consists of two closespaced dipoles or colinear arrays. Because of
the close spacing, it is possible to obtain the
Kraus
Beam
A
469
proper phase relationships in multi- section
flat tops by crossing the wires at the voltage
loops, rather than by resorting to phasing
stubs. This greatly simplifies the array. (See
figure 23.) Any number of sections may be
used, though the one- and two -section arrangements are the most popular. Little extra gain
is obtained by using more than four sections,
and trouble from phase shift may appear.
A center -fed single- section flat -top beam
cut according to the table, can be used quite
successfully on its second harmonic, the pattern being similar except that it is a little
sharper. The single- section array can also be
used on its fourth harmonic with some success,
though there then will be four cloverleaf lobes,
much the same as with a full -wave antenna.
If a flat -top beam is to be used on more than
one band, tuned feeders are necessary.
The radiation resistance of a flat-top beam
is rather low, especially when only one section is used. This means that the voltage will
be high at the voltage loops. For this reason,
especially good insulators should be used for
best results in wet weather.
The exact lengths for the radiating elements
are not especially critical, because slight deviations from the correct lengths can be compensated in the stub or tuned feeders. Proper
stub adjustment is covered in Chapter Twentyfive. Suitable radiator lengths and approximate
stub dimensions are given in the accompanying design table.
Figure 23 shows top views of eight types
of flat -top beam antennas. The dimensions for
using these antennas on different bands are
given in the design table. The 7- and 28 -Mc.
bands are divided into two parts, but the dimensions for either the low- or high -frequency
ends of these bands will be satisfactory for
use over the entire band.
In any case, the antennas are tuned to the
frequency used, by adjusting the shorting wire
on the stub, or tuning the feeders, if no stub
is used. The data in the table may be extended
to other bands or frequencies by applying the
proper factor. Thus, for 50 to 52 Mc. operation,
the values for 28 to 29 Mc. are divided by 1.8.
All of the antennas have a bidirectional horizontal pattern on their fundamental frequency.
The maximum signal is broadside to the flat
top. The single- section type has this pattern
on both its fundamental frequency and second
harmonic. The other types have four main lobes
of radiation on the second and higher harmonics. The nominal gains of the different types
over a half -wave comparison antenna are as
follows: single- section, 4 db; two- section, 6
db; four- section, 8 db.
The maximum spacings given make the
beams less critical in their adjustments. Up
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L
L
L
RADIO
THE
High Frequency Directive Antennas
468
DI
DI
D2
3
saw
END-LINK COIL
TO TUNE FREQUENCY
C.
100 LUF
DIMENSIONS
30011 RIBBON LINE
DIMENSIONS
IOM.
134.
20M.
GAIN APPROX. 7.5 Da
IT'
22,3 33.6"
tree 22.9- 34'6'
L
D
Figure
21
THE "SIX- SHOOTER" BROADSIDE ARRAY
wire line should be employed if the antenna is
used with a high power transmitter.
To tune the reflector, the back of the antenna is aimed at a nearby field -strength meter
and the reflector stub capacitor is adjusted
for minimum received signal at the operating
frequency.
This antenna provides high gain for its small
size, and is recommended for 28 -Mc. work.
The elements may be made of number 14 enamel wire, and the array may be built on a light
bamboo or wood framework.
The "Six- Shooter"
Broadside Array
As a good compromise between gain, directivity,
compactness, mechanical
simplicity, ease of adjustment, and band width
the array of figure 21 is recommended for the
10 to 30 Mc. range when the additional array
width and greater directivity are not obtainable. The free space directivity gain is approximately 7.5 db over one element, and the
practical dx signal gain over one element at
the same average elevation is of about the
same magnitude when the array is sufficiently
elevated. To show up to best advantage the
array should be elevated sufficiently to put
the lower elements well in the clear, and preferably at least 0.5 wavelength above ground.
Another application of vertical orientation of the raBroadside Curtain
diating elements of an array in order to obtain low angle radiation at the lower end of the h -f
range with low pole heights is illustrated in
figure 22. When precut to the specified dimensions this single pattern array will perform
well over the 7 -Mc. amateur band or the 4 -Mc.
amateur phone band. For the 4 -Mc. band the
required two poles need be only 70 feet high,
and the array will provide a practical signal
The
"Bobtail"
Bidirectional
32 a
COAXIAL LINE
DI
Dz
40M.
60M.
126,
33,
60,
D3 30.7036 34'TOM'
"BOBTAIL"
Figure 22
BIDIRECTIONAL
BROAD-
SIDE CURTAIN FOR THE 7 -MC. OR THE
4.0 -MC. AMATEUR BANDS
This simple vertically polarized array provides low angle radiation and response with
comparatively low pole heights, and is very
effective for dx work on the 7 -Mc. band or
the 4.0 -Mc. phone band. Because of the
phase relationships, radiation from the horizontal portion of the antenna is effectively
suppressed. Very little current flows in the
ground lead to the coupling tank; so an elaborate ground system is not required, and the
length of the ground lead is not critical so
long os it uses heavy wire and is reasonably short.
gain averaging from 7 to 10 db over a horizontal half-wave dipole utilizing the same pole
height when the path length exceeds 2500
miles.
The horizontal directivity is only moderate,
the beam width at the half power points being
slightly greater than that obtained from three
cophased vertical radiators fed with equal currents. This is explained by the fact that the
current in each of the two outer radiators of
this array carries only about half as much current as the center, driven element. While this
"binomial" current distribution suppresses
the end -fire lobe that occurs when an odd number of parallel radiators with half-wave spacing are fed equal currents, the array still exhibits some high -angle radiation and response
off the ends as a result of imperfect cancellation in the flat top portion. This is not sufficient to affect the power gain appreciably, but
does degrade the discrimination somewhat.
A moderate amount of sag can be tolerated
at the center of the flat top, where it connects
to the driven vertical element. The poles and
antenna tank should be so located with respect
to each other that the driven vertical element
drops approximately straight down from the
flat top.
www.americanradiohistory.com
HANDBOOK
Broadside Arrays
467
EACN SIOE
CDR
CAIN
sT
EACH
SIDE
REFLECTOR
C'
LLF
PUNK FOR M
MIN
PICKUP
OF BEI M.
NOTE
I
I
RM
REAR
'RADIATOR
5% 8.107
Id
SPACED
s,oe LENCTN- rI' B
IT T
)^
ON
BALANCED
CED LINE
TUNING UNIT OR
TRANSMITTER.
FOR 2I MC
FOR 74 MC.
ELEMENT SPACING PB- FOR LACS BAND.
STUD
LENCT PPROX IS'
20
DIMENSIONS
101.4
ISM.
FOR 21 MC.
FOR 14 MC.
50M.
B'B 2YY SS'
D B'S 12.57
O
SECTION
151E. WIRE
SPACED
4-
GAIN APPROX.
DB
150.11 LINE TO TRANSMITTER
THE
Figure
THE
Figure 20
CUBICAL -QUAD ANTENNA FOR
THE 10 -METER BAND
19
"BI- SQUARE" BROADSIDE ARRAY
This bidirectional array Is related to the
and in spite of the oblique elements, is horizontally polarized. It has slightly less gain and directivity than the Lazy
H, the free space directivity gain being approximately 4 db. Its chief advantage is the
fact that only a single pole is required for
"Lazy H,"
support, and two such arrays may be supported from a single pole without interaction
if the planes of the elements are at right
angles. A 600 -ohm line may be substituted
for the Twin-Lead, and either operated os a
resonant line, or made non -resonant by the
incorporation of a matching stub.
still worthwhile, being approximately 4 db over
a half-wave horizontal dipole at the same average elevation.
When two Bi -Square arrays are suspended
at right angles to each other (for general coverage) from a single pole, the Q sections
should be well separated or else symmetrically
arranged in the form of a square (the diagonal
conductors forming one Q section) in order to
minimize coupling between them. The same
applies to the line if open construction is used
instead of Twin -Lead, but if Twin -Lead is
used the coupling can be made negligible simply by separating the two Twin -Lead lines by
at least two inches and twisting one TwinLead so as to effect a transposition every foot
or so.
When tuned feeders are employed, the BiSquare array can be used on half frequency as
an end -fire vertically polarized array, giving
a slight practical dx signal gain over a vertical half -wave dipole at the same height.
A second Bi -Square serving as a reflector
may be placed 0.15 wavelength behind this antenna to provide an overall gain of 8.5 db. The
reflector may be tuned by means of a quarterwave stub which has a moveable shorting bar
at the bottom end. The stub is used as a substitute for the Q- section, since the reflector
employs no feed line.
smaller version of the BiSquare antenna is the Cubical -Quad antenna. Two halfwaves of wire are folded into a square that is
one -quarter wavelength on a side, as shown in
figure 20. The arraÿ radiates a horizontally
polarized signal. A reflector placed 0.15 wavelength behind the antenna provides an overall
gain of some 6 db. A shorted stub with a paralleled tuning capacitor is used to resonate the
The "CubicalQuad" Antenna
A
reflector.
The Cubical -Quad is fed with a 150-ohm
line, and should employ some sort of antenna
tuner at the transmitter end of the line if a pinetwork type transmitter is used. There is a
small standing wave on the line, and an open
www.americanradiohistory.com
THE
High Frequency Directive Antennas
466
D
L
+
RADIO
'
L
L
-
L
'
1111111111
GAIN APPROX. 5 DB
9004 LINE
DIMENSION L
10M
13'9-
DIMENSIONS
10M.
GAIN APPRO*. 3 DB
I5M. 20M.
L 13'3- 22' 32'10.
S 20'
30' 40'
P 14.2 21'3 213'4
D
3'Y
7.0-
X -ARRAY
7511 TRANSMISSION LINE
FOR
17
28 MC.,
21
17.3
ANTENNA
TUNER
TRANSMITTER
Figure 18
DOUBLE -BRUCE ARRAY FOR 10,
15, AND 20 METERS
If a 600 -ohm feed line is used, the 20 -meter
array will also perform on 10 meters as o
Sterba curtain, with an approximate gain of
9 db.
MC.,
OR 14 MC.
The entire array (with the exception of the
75 -ohm feed line) is constructed of 300 -ohm
ribbon line. Be sure phasing lines (P) are
poled correctly, as shown.
in a vertical plane and properly phased, a
simplified form of in -phase curtain is formed,
providing an overall gain of about 6 db. Such
an array is shown in figure 17. In this X- array,
the four dipoles are all in phase, and are fed
by four sections of 300 -ohm line, each one half wavelength long, the free ends of all four
lines being connected in parallel. The feed
impedance at the junction of these four lines
is about 75 ohms, and a length of 75 -ohm
Twin -Lead may be used for the feedline to
the array.
An array of this type is quite small for the
28 -Mc. band, and is not out of the question
for the 21 -Mc. band. For best results, the bottom section of the array should be one -half
wavelength above ground.
The Double -Bruce
Array
20 /IOM.
THE
Figure
THE
I5M.
113'
The Bruce Beam consists
of a long wire folded so
that vertical elements
carry in -phase currents while the horizontal
elements carry out of phase currents. Radiation from the horizontal sections is low since
only a small current flows in this part of the
wire, and it is largely phased-out. Since the
height of the Bruce Beam is only one -quarter
wavelength, the gain per linear foot of array
is quite low. Two Bruce Beams may be combined as shown in figure 18 to produce the
Double Bruce array. A four section Double
Bruce will give a vertically polarized emission, with a power gain of 5 db over a simple
dipole, and is a very simple beam to construct.
This antenna, like other so- called "broadside"
arrays, radiates maximum power at right angles
to the plane of the array.
The feed impedance of the Double Bruce is
about 750 ohms. The array may be fed with a
one -quarter wave stub made of 300 -ohm ribbon
line and a feedline made of 150 -ohm ribbon
line. Alternatively, the array may be fed directly with a wide- spaced 600 -ohm transmission line (figure 18). The feedline should
be brought away from the Double Bruce for a
short distance before it drops downward, to
prevent interaction between the feedline and
the lower part of the center phasing section of
the array. For best results, the bottom sections of the array should be one -half wavelength above ground.
Arrays such as the X -array and the Double
Bruce are essentially high impedance devices,
and exhibit relatively broad -band characteristics. They are less critical of adjustment than
a parasitic array, and they work well over a
wide frequency range such as is encountered
on the 28 -29.7 Mc. band.
The "Bi- Square"
Broadside Array
Illustrated in figure
19
is a
simple method of feeding a
small broadside array first
described by W6BCX several years ago as a
practical method of suspending an effective
array from a single pole. As two arrays of this
type can be supported at right angles from a
single pole without interaction, it offers a
solution to the problem of suspending two arrays in a restricted space with a minimum of
erection work. The free space directivity gain
is slightly less than that of a Lazy I1, but is
www.americanradiohistory.com
HANDBOOK
Broadside Arrays
465
NON - RESONANT FELDER
GAIN APPROX. 6 DB
GAIN APPROX. 8 DB
Figure 16
THE STERBA CURTAIN ARRAY
Approximate directive gains along with alternative feed methods are shown.
GAIN APPROX. 8 DB
sent points of maximum current. All arrows
should point in the same direction in each portion of the radiating sections of an antenna in
order to provide a field in phase for broadside
radiation. This condition is satisfied for the
arrays illustrated in figure 16. Figures 16A
and 16C show simple methods of feeding
a short Sterba curtain, while an alternative
method of feed is shown in the higher gain antenna of figure 16B.
In the case of each of the arrays of figure
16, and also the "Lazy H" of figure 15, the
array may be made unidirectional and the gain
increased by 3 db if an exactly similar array
is constructed and placed approximately 14
wave behind the driven array. A screen or mesh
of wires slightly greater in area than the antenna array may be used instead of an additional array as a reflector to obtain a unidirectional system. The spacing between the reflecting wires may vary from 0.05 to 0.1 wavelength with the spacing between the reflecting
wires the smallest directly behind the driven
elements. The wires in the untuned reflecting
system should be parallel to the radiating elements of the array, and the spacing of the complete reflector system should be approximately
0.2 to 0.25 wavelength behind the driven elements.
On frequencies below perhaps 100 Mc. it
normally will be impracticable to use a wire screen reflector behind an antenna array such
as a Sterba curtain or a "Lazy H." Parasitic
elements may be used as reflectors or directors, but parasitic elements have the disadvantage that their operation is selective with respect to relatively small changes in frequency.
Nevertheless, parasitic reflectors for such arrays are quite widely used.
In section 23 -5 it was shown
how two dipoles may be arranged
in phase to provide a power gain of (some) 3
db. If two such pairs of dipoles are stacked
The
X -Array
LAZY -H AND STERBA
(STACKED DIPOLE) DESIGN TABLE
FREQUENCY
IN MC.
7.0
7.3
14.0
14.2
14.4
21.0
21.5
27.3
28.0
29.0
50.0
52.0
54.0
144.0
146.0
148.0
www.americanradiohistory.com
L,
68'2"
65'10"
34'1"
33'8"
33'4"
22'9"
22'3"
17'7"
17'
16'6"
9'7"
9'3"
8'10"
39.8"
39"
38.4"
L.
70'
67'6"
35'
34.7"
34'2"
23'3"
22'9"
17'10"
17'7"
17'
9'10"
9'S"
9'1"
40.5"
40"
39.5"
L,
35'
33'9"
17'6"
17'3"
17'
11'B"
11'5"
8'11"
8'9"
8'6"
4'11"
4'8"
4'6"
20.3'
20"
19.8"
464
THE
High Frequency Directive Antennas
of a colinear antenna is proportional to the
RADIO
Li
LI
overall length, whether the individual radiating
elements are 1/4 wave, 1/2 wave or 1/4 wave in
length.
The gain of two colinear half
waves may be increased by
increasing the physical spacing between the elements, up to a maximum of
about one half wavelength. If the half wave
elements are fed with equal lengths of transmission line, poled correctly, a gain of about
3.3 db is produced. Such an antenna is shown
in figure 13. By means of a phase reversing
switch, the two elements may be operated out
of phase, producing a cloverleaf pattern with
slightly less maximum gain.
A three element "precut" array for 40 meter
operation is shown in figure 14. It is fed directly with 300 ohm "ribbon line," and may
be matched to a 52 ohm coaxial output transmitter by means of a Balun, such as the Barker
&
illiamson 3975. The antenna has a gain of
about 3.2 db, and a beam width at half -power
points of 40 degrees.
Spaced Half
Wave Antennas
23 -6
OUARTER-WAVE STUR
NON -RESONANT
FEED LIN
CAIN APPROX. 5.5 DR
Broadside Arrays
Colinear elements may be stacked above or
below another string of colinear elements to
produce what is commonly called a broadside
array. Such an array, when horizontal elements
are used, possesses vertical directivity in
proportion to the number of broadsided (vertically stacked) sections which have been
used. Since broadside arrays do have good vertical directivity their use is recommended on
the 14 -Mc. band and on those higher in frequency. One of the most popular of simple
broadside arrays is the "Lazy 11" array of figure 15. Horizontal colinear elements stacked
two above two make up this antenna system
which is highly recommended for work on frequencies above perhaps 14 -Mc. when moderate
gain without too much directivity is desired.
It has high radiation resistance and a gain of
approximately 5.5 db. The high radiation resistance results in low voltages and a broad
resonance curve, which permits use of inexpensive insulators and enables the array to be
used over a fairly wide range in frequency.
For dimensions, see the stacked dipole design
table.
Vertical stacking may be applied
Dipoles
to strings of colinear elements
longer than two half waves. In
such arrays, the end quarter wave of each
string of radiators usually is bent in to meet
Stacked
2
RESONANT FEED LINE
Figure 15
THE "LAZY H" ANTENNA SYSTEM
Stacking the colinear pairs gives both horizontal and vertical directivity. As shown, the
array will give about 5.5 db gain. Note that
the array may be fed either at the center of
the phasing section or at the bottom; if fed
at the bottom the phasing section must be
twisted through 180 °.
a similar bent quarter wave from the opposite
end radiator. This provides better balance and
better coupling between the upper and lower
elements when the array is current -fed. Arrays
of this type are shown in figure 16, and are
commonly known as curtain arrays.
Correct length for the elements and stubs
can be determined for any stacked dipole array
from the Stacked -Dipole Design Table.
In the sketches of figure 16 the arrowheads
represent the direction of current flow at any
given instant. The dots on the radiators repre-
www.americanradiohistory.com
Colinear Arrays
HANDBOOK
COLINEAR ANTENNA DESIGN CHART
FREQUENCY
IN MC.
Li
La
16'8'
22'e"
14.2
.0
L3
e'6
17'
1+'e
33'e
23'3
34.7
e7'
66'6"
34'4
61'6
68'2
120'
133'
3.e
RuCI
FMC)
s
26.5
21.2
7.15
463
F(I)
A B
A-B
=15011 FEED POINT
GAINAPPROX. 3D6
17'3
123'
136'5-
Figure 12
DOUBLE EXTENDED ZEPP ANTENNA
For best results, antenna should be tuned to
operating frequency by means of griddip
oscillator.
simple colinear antenna array
very effective radiating system
for the 3.5 -Mc. and 7.0 -Mc. bands,
but its use is not recommended on higher frequencies since such arrays do not possess
any vertical directivity. The elevation radiation pattern for such an array is essentially
the same as for a half-wave dipole. This consideration applies whether the elements are of
normal length or are extended.
The colinear antenna consists of two or
more radiating sections from 0.5 to 0.65 wavelengths long, with the current in phase in each
section. The necessary phase reversal between
sections is obtained through the use of resonant tuning stubs as illustrated in figure 11.
The gain of a colinear array using half -wave
elements (in decibels) is approximately equal
to the number of elements in the array. The
exact figures are as follows:
2
3
4
6
Number of Elements
5
Gain in Decibels
1.8 3.3 4.5 5.3 6.2
As additional in -phase colinear elements
are added to a doublet, the radiation resistance
goes up much faster than when additional half
waves are added out of phase (harmonic operated antenna).
For a colinear array of from 2 to 6 elements,
Colinear
Arrays
The
is
h~- e'(,`1
a
i'
F(1AC)
the terminal radiation resistance in ohms at
any current loop is approximately 100 times
the number of elements.
It should be borne in mind that the gain from
a colinear antenna depends upon the sharpness
of the horizontal directivity since no vertical
directivity is provided. An array with several
colinear elements will give considerable gain,
but will have a sharp horizontal radiation
pattern.
The gain of a conventional
two- element Franklin colinear antenna can be increased
to a value approaching that obtained from a
three -element Franklin, simply by making the
two radiating elements 230° long instead of
180° long. The phasing stub is shortened correspondingly to maintain the whole array in
resonance. Thus, instead of having 0.5 -wavelength elements and 0.25-wavelength stub, the
elements are made 0.64 wavelength long and
the s tub is approximately 0.11 wavelength
Double Extended
Zepp
long.
Dimensions for the double extended Zepp
are given in figure 12.
The vertical directivity of a colinear antenna having 230° elements is the same as for
one having 180° elements. There is little advantage in using extended sections when the
total length of the array is to be greater than
about 1.5 wavelength overall since the gain
r'--
65 e
-Ai
MID
32'9
PHASE -REVERSING SWITCH
FOR CLOVERLEAF PATTERN
MAKE STUBS
65 6
-H
TOeE.
14
65
e-+{
e ms
32'9'
-
OFl
WIRE,SPACED
-
"-3OOR
RIBBON TO
TRANSMITTER, ANY LENGTH
GAIN APPROX. 3 DB
Figure
13
Figure
14
TWO COLINEAR HALF -WAVE ANTENNAS
IN PHASE PRODUCE A 3 DB GAIN WHEN
PRE -CUT LINEAR ARRAY FOR 40 -METER
SEPARATED ONE -HALF WAVELENGTH
OPERATION
www.americanradiohistory.com
High Frequency Directive Antennas
462
---Lt
f
4-
L2
THE
RADIO
L2
Lt
-----e--
-
_
-
L!
PLANE OF WIRES
END VIEW
L3
QUARTER-WAVE STUBS
NON- RESONANT
FEED LIN
14)
5=
=
ISO. OUT OF PHASE
(FLAT -TOP BEAM, ETC.)
I
/
IN PHASE
(LAZY H, SIERRA CURTAIN)
Figure 10
RADIATION PATTERNS OF A PAIR OF
DIPOLES OPERATING WITH IN -PHASE
EXCITATION, AND WITH EXCITATION
GAIN APPROS
Figure
4
S
DB
11
THE FRANKLIN OR COLINEAR
ANTENNA ARRAY
An antenna of this type, regardless of the
number of elements, attains all of its directivity through sharpening of the horizontal
or azimuth radiation pattern; no vertical directivity is provided. Hence a long antenna
of this type has an extremely sharp azimuth
pattern, but no vertical directivity.
180° OUT OF PHASE
If the dipoles are oriented horizontally most
of the directivity will be in the vertical
plane; if they are oriented vertically most
of the directivity will be in the horizontal
plane.
and 180° (45 °, 90 °, and 135° for instance),
the pattern is unsymmetrical, the radiation be-
ing greater in one direction than in the oppo-
site direction.
With spacings of more than 0.8 wavelength,
more than two main lobes appear for all phasing combinations; hence, such spacings are
seldom used.
With the dipoles driven so as to
be in phase, the most effective
spacing is between 0.5 and 0.7
wavelength. The latter provides greater gain,
but minor lobes are present which do not appear at 0.5- wavelength spacing. The radiation
is broadside to the plane of the wires, and the
gain is slightly greater than can be obtained
from two dipoles out of phase. The gain falls
off rapidly for spacings less than 0.375 wavelength, and there is little point in using spacing of 0.25 wavelength or less with in -phase
dipoles, except where it is desirable to increase the radiation resistance. (See Multi In -Phase
Spacing
Wire
Doublet.)
dipoles are fed 180°
out of phase, the directivity is
through the plane of the wires,
and is greatest with close spacing, though
there is but
difference in the pattern
after the spacing is made less than 0.125
wavelength. The radiation resistance becomes
so low for spacings of less than 0.1 wavelength that such spacings are not practicable.
Out of Phase
Spacing
When the
little
In the three foregoing examples, most of the
a plane at a right
angle to the wires, though when out of phase,
the directivity is in a line through the wires,
and when in phase, the directivity is broadside
to them. Thus, if the wires are oriented verti-
directivity provided is in
cally, mostly horizontal directivity will be
provided. If the wires are oriented horizontally,
most of the directivity obtained will be vertical directivity.
To increase the sharpness of the directivity
in all planes that include one of the wires,
additional identical elements are added in the
line of the wires, and fed so as to be in phase.
The familiar H array is one array utilizing
both types of directivity in the manner prescribed. The two -section Kraus flat -top beam
is another.
These two antennas in their various forms
are directional in a horizontal plane, in addition to being low -angle radiators, and are perhaps the most practicable of the bidirectional
stacked -dipole arrays for amateur use. More
phased elements can be used to provide greater directivity in planes including one of the
radiating elements. The fl then becomes a
Sterba- curtain array.
For unidirectional work the most practicable
stacked -dipole arrays for amateur -band use
are parasitically- excited systems using relatively close spacing between the reflectors
and the directors. Antennas of this type are
described in detail in a later chapter. The
next most practicable unidirectional array is
an H or a Sterba curtain with a similar system
placed approximately one -quarter wave behind.
The use of a reflector system in conjunction
with any type of stacked -dipole broadside array will increase the gain by 3 db.
www.americanradiohistory.com
HANDBOOK
Antenna
Rhombic
The
461
J,
Figure
8
TYPICAL RHOMBIC
ANTENNA DESIGN
The antenna system illusabove may be used
over the frequency ronge from
7 to 29 Mc. without change.
The directivity of the system
may be reversed by the system discussed in the text.
trated
LINE TO TX
N14
SPACED
e'
S. 214 FEET
SPACING BETWEEN SIDES
TOTAL LENGTH
TERMINATING LINE
OF 250' OF N 26
NICHROME SPACED 6"
AND B00 -OHM 16 -WATT
5112 FEET
H50
This antenna will give about 11 db
gain in the 14.0 -Mc. band. The approximate
gain of a rhombic antenna over a dipole, both
above normal soil, is given in figure 9.
A considerable amount of directivity is lost
when the terminating resistor is left off the
end and the system is operated as a resonant
antenna. If it is desired to reverse the direction of the antenna it is much better practice
to run transmission lines to both ends of the
antenna, and then run the terminating line to
the operating position. Then with the aid of
two d -p -d -t switches it will be possible to connect either feeder to the antenna changeover
switch and the other feeder to the terminating
line, thus reversing the direction of the array
and maintaining the same termination for
bands.
Stacked -Dipole
23 -5
Arrays
The characteristics of a half -wave dipole
already have been described. When another
dipole is placed in the vicinity and excited
either direction of operation.
Figure 7 gives curves for optimum- design
rhombic antennas by both the maximum-output method and the alignment method. The
alignment method is about 1.5 db down from
the maximum output method but requires only
about 0.74 as much leg length. The height and
tilt angle is the same in either case. Figure
8 gives construction data for a recommended
rhombic antenna for the 7.0 through 29.7 Mc.
either directly or parasitically, the resultant
radiation pattern will depend upon the spacing
and phase differential, as well as the relative
magnitude of the currents. With spacings less
than 0.65 wavelength, the radiation is mainly
broadside to the two wires (bidirectional) when
there is no phase difference, and through the
wires (end fire) when the wires are 180° out
of phase. With phase differences between 0°
ILI
J
ie
215
p 14
w
Figure 9
RHOMBIC ANTENNA GAIN
Showing the theoretical gain of a rhombic
antenna, in terms of the side length, over a
half -wave antenna mounted at the same
height above the same type of soil.
CARBON RESISTOR AT
6 2-WATT 100-OHM
END
RESISTORS IN SERIES
1
312
LL
11
3
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2
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6
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10
11
R
13 14 15
le Ti
16
/f
20
OF EACH LEG OF RHOMBIC IN WAVELENGTHS
www.americanradiohistory.com
460
23-4
The Rhombic
Antenna
....
....
Rhombic
Termination
H
RADIO
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The terminated rhombic or diamond is probably the most effective directional antenna
that is practical for amateur communication.
This antenna is non -resonant, with the result
that it can be used on three amateur bands,
such as 10, 20, and 40 meters. When the antenna is non -resonant, i.e., properly terminated, the system is undirectional, and the
wire dimensions are not critical.
.
.
.
.
o
is terminated
resistance of a value
When the free end
with
a
between 700 and 800 ohms the
backwave is eliminated, the forward gain is
increased, and the antenna can be used on
several bands without changes. The terminating resistance should be capable of dissipating one -third the power output of the transmitter, and should have very little reactance.
For medium or low power transmitters, the
non -inductive plaque resistors will serve as
a satisfactory termination. Several manufacturers offer special resistors suitable for terminating a rhombic antenna. The terminating device should, for technical reasons, present a
small amount of inductive reactance at the
point of termination.
A compromise terminating device commonly
used consists of a terminated 250 -foot or
longer length of line, made of resistance wire
which does not have too much resistance per
unit length. If the latter qualification is not
met, the reactance of the line will be excessive. A 250 -foot line consisting of no. 25
nichrome wire, spaced 6 inches and terminated
with 800 ohms, will serve satisfactorily. Because of the attenuation of the line, the
lumped resistance at the end of the line need
dissipate but a few watts even when high power is used. A half-dozen 5000 -ohm 2 -watt carbon resistors in parallel will serve for all except very high power. The attenuating line
may be folded back on itself to take up less
room.
The determination of the best value of terminating resistor may be made while receiving,
if the input impedance of the receiver is approximately 800 ohms. The value of resistor
which gives the best directivity on reception
will not give the most gain when transmitting,
but there will be little difference between the
two conditions.
The input resistance of the rhombic which
is reflected into the transmission line that
feeds it is always somewhat less than the
terminating resistance, and is around 700 to
750 ohms when the terminating resistor is 800
ohms.
THE
High Frequency Directive Antennas
r
r
M
Ir
n.
WAVE ANGLE
Figure
A
sr
ar
Zr
7
RHOMBIC ANTENNA DESIGN TABLE
Design data is given in terms of the wave
angle (vertical angle of transmission and reception) of the antenna. The lengths I are
for the "maximum output" design; the shorter
are for the "alignment" method
lengths
which gives approximately 1.5 db less gain
with o considerable reduction in the space
required for the antenna. The values of side
length, tilt angle, and height for a given
wave angle are obtained by drawing o vertical line upward from the desired wave
angle.
I'
The antenna should be fed with a non -resonant line having a characteristic impedance
of 650 to 700 ohms. The four corners of the
rhombic should be at least one -half wavelength above ground for the lowest frequency
of operation. For three-band operation the
proper tilt angle ,;4 for the center band should
be observed.
The rhombic antenna transmits
tally- polarized wave at
a
a
horizon-
relatively low angle
above the horizon. The angle of radiation
(wave angle) decreases as the height above
ground is increased in the same manner as
with a dipole antenna. The rhombic should
not be tilted in any plane. In other words, the
poles should all be of the same height and
the plane of the antenna should be parallel
with the ground.
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HANDBOOK
I/l
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I
If
1:/
1P.i
>lo11SL
The
Antenna
V
459
13
3
w.
'
b
TRAN3MIT
RECEIVE
t
yl
o
Jul
u
p
Figure
TYPICAL
"V"
Z
z
5
BEAM ANTENNA
M16
411(Ii
7
3=1111/411011111
2
RIMIIIIIME111101111111.11MMIEll
oo
2
2
4
3
6
LENGTH OF SIDE
7
"L"
other removes two of the four main lobes, and
increases the other two in such a way as to
form two lobes of still greater magnitude.
The correct wire lengths and the degree of
the angle b are listed in the V- Antenna Design
Table for various frequencies in the 10 -, 20and 40 -meter amateur bands. Apex angles for
all side lengths are given in figure 4. The
gain of a "V" beam in terms of the side length
when optimum apex angle is used is given in
figure 6.
The legs of a very long V antenna are usually so arranged that the included angle is twice
the angle of the major lobe from a single wire
if used alone. This arrangement concentrates
the radiation of each wire along the bisector
of the angle, and permits part of the other
lobes to cancel each other.
üith legs shorter than 3 wavelengths, the
best directivity and gain are obtained with a
somewhat smaller angle than that determined
by the lobes. Optimum directivity for a one wave V is obtained when the angle is 90°
rather than 180 °, as determined by the ground
pattern alone.
If very long wires are used in the V, the
angle between the wires is almost unchanged
when the length of the wires in wavelengths
is altered. However, an error of a few degrees
causes a much larger loss in directivity and
gain in the case of the longer V than in the
shorter one.
The vertical angle at which the wave is
best transmitted or received from a horizontal
V antenna depends largely upon the included
angle. The sides of the V antenna should be
at least a half wavelength above ground; commercial practice dictates a height of approximately a full wavelength above ground.
=2r
6° 70
L
6'so
26000
29000
34'6"
69'6
33.6
21100
L
=4a
L =BT
6
=52
6F39
260'
67.3"
140'
135'
45'9"
91.9"
163'
21300
45'4"
91'4"
162'6
366'
365'
14050
14150
14250
69'
7020
7100
7200
66'6"
66'2'
136'2'
136.6'
134.10"
12
ground, in terms of the side length L.
V- ANTENNA DESIGN TABLE
L= ñ
11
Figure 6
DIRECTIVE GAIN OF A "V" BEAM
This curve shows the approximate directive
gain of a V beam with respect to a half-wave
antenna located the same distance above
for a long wire. The reaction of one upon the
FREQUENCY
IN KILOCYCLES
10
271'
139'
136'
137'
279'
277'
275'
356'
555'
276'
275'
271'
556'
1120'
1106'
1060'
www.americanradiohistory.com
552'
545'
552'
High Frequency Directive Antennas
458
THE
RADIO
4A
4ZA
LONG- ANTENNA DESIGN TABLE
APPROXIMATE LENGTH IN FEET-END-FED ANTENNAS
FREQUENCY
IN M
IZA
1A
3A
84
67
101
118
104
136 1/2
91 3/4
114 1/2
114 3/4
92
115
137
171
29
26
50
52
67
69
21.4
21.2
21.0
45
45 1M
66
66 1/4
911/2
66 1/2
14.2
14.0
67 1/2
88 1/2
7.3
7.15
7.0
.0
3.6
3.6
3.5
2.0
1.9
1.6
451/2
102
2
I39
103 1/2
174
138
136 1/2
137
206
276
207
207 1/2
277
277 1/2
346
347
348
240
232
362
465
616
361
511
60
268
403
414
540
555
676
696
1230
1280
274
480
304
725
972
763
1020
532
605
1060
152
122
135
140
160 1/2
160 3.4
165 1/2
163 3/4
2091/2
137
161
166
206
209
240
244
275
279
310
418
66
555
625
557
627
416
467
488
356
628
633
977
1030
1090
1120
1100
1160
136
3.41
730
770
612
633
900
950
977
157
209 3/4
210
31
1220
1473
form of a V, it is possible to make two of the
maximum lobes of one leg shoot in the same
direction as two of the maximum lobes of the
other leg of the V. The resulting antenna is
bidirectional (two opposite directions) for the
main lobes of radiation. Each side of the V
can be made any odd or even number of quarter
wavelengths, depending on the method of feeding the apex of the V. The complete system
must be a multiple of half waves. If each leg
is an even number of quarter waves long, the
antenna must be voltage -fed at the apex; if an
odd number of quarter waves long, current feed
must be used.
By choosing the proper apex angle, figure
and figure 5, the lobes of radiation from the
two long -wire antennas aid each other to form
a bidirectional beam. Each wire by itself
would have a radiation pattern similar to that
4
The V Antenna
If
3
17
One of the most practical methods of feeding a long -wire antenna is to bring one end of
it into the radio room for direct connection to
a tuned antenna circuit which is link- coupled
through a harmonic- attenuating filter to the
transmitter. The antenna can be tuned effectively to resonance for operation on any harmonic by means of the tuned circuit which is
connected to the end of the antenna. A ground
is sometimes connected to the center of the
tuned coil.
If desired, the antenna can be opened and
current -fed at a point of maximum current b'
means cf low- impedance ribbon line, or by a
quarter -wave matching section and open line.
23 -3
X
+A
2X
33
34
two long -wire antennas are built in the
ISO
140
Figure 4
INCLUDED ANGLE FOR A
120
"V"
BEAM
Showing the included angle between the legs of a V beam for
various leg lengths. For optimum alignment of the radiation
lobe at the correct vertical
angle with leg lengths less thon
three wavelengths, the optimum
Included angle is shown by the
dashed curve.
40
20
o
o
4
LENGTH IN
10
6
"L'
12
WAVELENGTHS
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HANDBOOK
Long
Wire
Radiators
457
LONG STRAIGHT WIRE ANTENNAS
Figure 3
DIRECTIVE GAIN OF
LONG -WIRE ANTENNAS
2
°o
2
3
4
3
7
e
s
10
DB POWER RATIO OF MAIN LOBE TO A DIPOLE
Types of
There is an enormous variety of directive antenna arrays that can give a substantial power gain in the desired direction of transmission or reception. However, some are more
effective than others requiring the same space.
In general it may be stated that long -wire antennas of various types, such as the single
long wire, the V beam, and the rhombic, are
less effective for a given space than arrays
composed of resonant elements, but the long wire arrays have the significant advantage
that they may be used over a relatively large
frequency range while resonant arrays are usable only over a quite narrow frequency band.
The horizontal radiation pattern of such antennas depends upon the vertical angle of radiation being considered. If the wire is more
than 4 wavelengths long, the maximum radiation at vertical angles of 15° to 20° (useful
for dx) is in line with the wire, being slightly
greater a few degrees either side of the wire
than directly off the ends. The directivity of
the main lobes of radiation is not particularly
sharp, and the minor lobes fill in between the
main lobes to permit working stations in nearly all directions, though the power radiated
broadside to the radiator will not be great if
the radiator is more than a few wavelengths
long. The directive gain of long -wire antennas,
in terms of the wire length in wavelengths is
given in figure 3.
Long Wire Radiators
To maintain the out -of-phase condition in
adjoining half -wave elements throughout the
length of the radiator, it is necessary that a
harmonic antenna be fed either at one end or
at a current loop. If fed at a voltage loop, the
adjacent sections will be fed in phase, and a
different radiation pattern will result.
The directivity of a long wire does not increase very much as the length is increased
beyond about 15 wavelengths. This is due to
the fact that all long -wire antennas are adversely affected by the r -f resistance of the
wire, and because the current amplitude begins to become unequal at different current
loops, as a result of attenuation along the wire
caused by radiation and losses. As the length
is increased, the tuning of the antenna becomes quite broad. In fact, a long wire about
15 waves long is practically aperiodic, and
works almost equally well over a wide range
of frequencies.
Directive Arrays
23 -2
Harmonically operated long wires radiate
better in certain directions than others, but
cannot be considered as having appreciable
directivity unless several wavelengths long.
The current in adjoining half -wave elements
flows in opposite directions at any instant,
and thus, the radiation from the various elements adds in certain directions and cancels
in others.
A half -wave do u b l e t in free space has a
"doughnut" of radiation surrounding it. A full
wave has 2 lobes, 3 half waves 3, etc. When
the radiator is made more than 4 half wavelengths long, the end lobes (cones of radiation) begin to show noticeable power gain over
a half-wave doublet, while the broadside lobes
get smaller and smaller in amplitude, even
though numerous (figure 2).
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456
High Frequency Directive Antennas
0.
M11Ì11111
7,
,,.
111111
,°. U1111III
,O.
S0.
20
,O
n.
1111111
-
1111
DOUBLE HOP
.
h.
SINGLE HOP
1111111
30
SO
100
1111
11111
111111
1101
111111
11111
__
_eÌ
1111
111111
HID
1111h
1111.,11
1
1
1
300 500
UMW
,000
3000
ro 000
GREAT CIRCLE DISTANCE IN MILES
Figure
1
OPTIMUM ANGLE OF RADIATION
WITH RESPECT TO DISTANCES
Shown above is o plot of the optimum angle
of radiation for one -hop and two -hop communication. An operating frequency close to
the optimum working frequency for the communication distance is assumed.
a directive antenna than to increase transmitter power, if more than a few watts of power
is being used.
Directive antennas for the high- frequency
range have been designed and used commercially with gains as high as 23 db over a simple dipole radiator. Gains as high as 35 db are
common in direct -ray microwave communication
and radar systems. A gain of 23 db represents
a power gain of 200 times and a gain of 35 db
represents a power gain of almost 3500 times.
However, an antenna with a gain of only 15 to
20 db is so sharp in its radiation pattern that
it is usable to full advantage only for point to -point work.
The increase in radiated power in the desired direction is obtained at the expense of
radiation in the undesired directions. Power
gains of 3 to 12 db seem to be most practicable for amateur communication, since the
width of a beam with this order of power gain
is wide enough to sweep a fairly large area.
Gains of 3 to 12 db represent effective transmitter power increases from 2 to 16 times.
use
There is a certain optimum
vertical angle of radiation
for sky -wave communication, this angle being dependent upon distance,
frequency, time of day, etc. Energy radiated at
an angle much lower than this optimum angle
is largely lost, while radiation at angles much
Horizontal Pattern
vs. Vertical Angle
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THE
RADIO
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HALT WAVE ANT.
--711G.++..-,
FULL WAVE ANT.
2 WAVES ANT.
HORIZONTAL ANTENNAS IN FREE SPACE
Figure 2
FREE -SPACE FIELD PATTERNS OF
LONG -WIRE ANTENNAS
The presence of the earth distorts the field
pattern in such a manner that the azimuth
pattern becomes a function of the elevation
angle.
higher than this optimum angle oftentimes is
not nearly so effective.
For this reason, the horizontal directivity
pattern as measured on the ground is of no import when dealing with frequencies and distances dependent upon sky -wave propagation.
It is the horizontal directivity (or gain or discrimination) measured at the most useful vertical angles of radiation that is of consequence. The horizontal radiation pattern, as
measured on the ground, is considerably different from the pattern obtained at a vertical
angle of 15 °, and still more different from a
pattern obtained at a vertical angle of 30 °.
In general, the energy which is radiated at
angles higher than approximately 30° above
the earth is effective at any frequency only for
local work.
For operation at frequencies in the vicinity
of 14 Mc., the most effective angle of radiation
is usually about 15° above the horizon, from
any kind of antenna. The most effective angles
for 10 -meter operation are those in the vicinity
of 10 °. Figure 1 is a chart giving the optimum
vertical angle of radiation for sky -wave propagation in terms of the great -circle distance between the transmitting and receiving antennas.
www.americanradiohistory.com
CHAPTER TWENTY -THREE
High
Frequency
Antenna Arrays
It is becoming of increasing importance in
most types of radio communication to be capable of concentrating the radiated signal from
the transmitter in a certain desired direction
and to be able to discriminate at the receiver
against reception from directions other than
the desired one. Such capabilities involve the
use of directive antenna arrays.
Few simple antennas, except the single vertical element, radiate energy equally well in
all azimuth (horizontal or compass) directions.
justed. They all are dipoles, and the feeder
system, if it does not radiate in itself, will
have no effect on the radiation pattern.
23 -1
Directive Antennas
When a multiplicity of radiating elements is
located and phased so as to reinforce the radiation in certain desired directions and to neutralize radiation in other directions, a directive antenna array is formed.
The function of a directive antenna when
used for transmitting is to give an increase in
signal strength in some direction at the expense of radiation in other directions. For reception, one might find useful an antenna giving little or no gain in the direction from which
it is desired to receive signals if the antenna
is able to discriminate against interfering signals and static arriving from other directions.
A good directive transmitting antenna, however,
can also be used to good advantage for reception.
If radiation can be confined to a narrow
beam, the signal intensity can be increased a
great many times in the desired direction of
transmission. This is equivalent to increasing
the power output of the transmitter. On the
higher frequencies, it is more economical to
All horizontal
antennas, except those specifically designed to give an omnidirectional azimuth radiation pattern such as the turnstile,
have some directive properties. These properties depend upon the length of the antenna in
wavelengths, the height above ground, and the
slope of the radiator.
The various forms of the half -wave horizontal antenna produce maximum radiation at right
angles to the wire, but the directional effect
is not great. Nearby objects also minimize the
directivity of a dipole radiator, so that it hardly seems worth while to go to the trouble to
rotate a simple half -wave dipole in an attempt
to improve transmission and reception in any
direction.
The half -wave doublet, folded dipole, zepp,
single- wire -fed, matched impedance, and Johnson Q antennas all have practically the same
radiation pattern when properly built and ad-
455
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Figure 45
REAR VIEW OF
TUNER SHOWING
PLACEMENT OF
MAJOR COMPON-
ENTS
Rotary inductor is drivby Johnson 116.2084 counter dial. Coaxial
receptacle JI
Input
Is mounted directly below rotary inductor.
n
termination. The transmitter is turned on
(preferably at reduced input) and resonance is
established in the amplifier tank circuit. The
sensitivity control of the tuner is adjusted to
provide near full scale deflection on the bridge
meter. Various settings of Si, L2, and Cl
should be tried to obtain a reduction of bridge
reading. As tuner resonance is approached,
the meter reading will decrease and the sensitivity control should be advanced. When the
system is in resonance, the meter will read
zero. All loading adjustments may then be
made with the transmitter controls. The tuner
should be readjusted whenever the frequency
of the transmitter is varied by an appreciable
amount.
ohm
Figure 46
CLOSE -UP
OF SWR BRIDGE
Simple SWR bridge is mounted
below the chassis of the tuner.
Carbon resistors are mounted to
two copper rings to form low
inductance
resistor.
one -ohm
Bridge capacitors form triangular
configuration
for lowest lead
inductance. Balancing capacitor
C2 is at lower right.
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HANDBOOK
Single -wire
Antenna
Tuner
453
52 a INPUT
FROM XMTR
R1
n
5
250
25
C2
1Q
5
SINGLE
TUNE
C1
330
2RV.
SENSITIVITY
Figure 43
ANTENNA TUNER IS HOUSED
IN METAL CABINET 7' x 8" IN
SIZE.
TAP AT
15 T., 27 T.,
FROM POINT A
INDUCTOR (10 NH)
Inductance switch SI and sensitivity control are at left with
(clockwise) position. The bridge is
balanced when the input impedance of the
tuner is 52 ohms resistive. This is the condition for maximum energy transfer between
transmission line and antenna. The meter is
graduated in arbitrary units, since actual SWR
value is not required.
Tuner
Construction
parts placement in
tuner is shown in
figures 43 and 45. Tapped
coil L1 is mounted upon 1-inch ceramic insulators, and all major components are mounted
above deck with the exception of the SWR
bridge (figure 46). The components of the
bridge are placed below deck, adjacent to
the coaxial input plug mounted on the rear
apron of the chassis. The ten 10 -ohm resistors
are soldered to two 1 -inch rings made of copper
wire as shown in the photograph. The bridge
capacitors are attached to this assembly with
extremely short leads.The 1N56 crystal mounts
at right angles to the resistors to insure minimum amount of capacitive coupling between the
resistors and the detector. The output lead
from the bridge passes through a ceramic feed thru insulator to the top side of the chassis.
Connection to the antenna is made by means
of a large feedthru insulator mounted on the
back of the tuner cabinet. This insulator is
not visible in the photographs.
Major
the
VARIABL E
CI- JOHNSON
350E20
C2- CENTRALA8
J1 -TYPE SO -239
TYPE 822
RECEPTACLE
R1-TEN 10-OHM -WATT CAR1
BON RESISTORS IN
PARA-
LLEL. INC TYPE LTA
Figure 44
SCHEMATIC, SINGLE -WIRE
ANTENNA TUNER
counter dial for L2 at center.
Output tuning capacitor CI is at
right.
SWR
meter is mounted
above SI.
mum
'll
=
-1
L1- 35 TURNS e 18, 2- DIA.,
3.9- LONG (A /R -DL/.e)
L2- JOHNSON 229 -207
ANT
010V
MICA
0
WIRE
L2
Si
Bridge
The SWR bridge must be
calibrated for 52 ohm service. This can be done by
temporarily disconnecting the lead between the
bridge and the antenna tuner and connecting a
2 -watt, 52 ohm carbon resistor to the junction
of R1 and the negative terminal of the 1N56
diode. The opposite lead of the carbon resistor
is grounded to the chassis of the bridge. A
small amount of r-f energy is fed to the input
of the bridge until a reading is obtained on
the r -f voltmeter. The 25 mmfd bridge balancing
capacitor C2 (see figure 46) is then adjusted
with a fibre -blade screwdriver until a zero
reading is obtained on the meter. The sensitivity control is advanced as the meter null
grows, in order to obtain the exact point of
bridge balance. When this point is found, the
carbon resistor should be removed and the
bridge attached to the antenna tuner. The
bridge capacitor is sealed with a drop of nail
polish to prevent misadjustment.
Calibration
All tuning adjustments are
made to obtain proper
transmitter loading with a
balanced (zero meter reading) bridge condition.
The tuner is connected to the transmitter
through a random length of 52 ohm coaxial
line, and the single wire antenna is attached
to the output terminal of the tuner. Transmitter
loading controls are set to approximate a 52
Tuner
Adjustments
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THE
Antennas and Antenna Matching
452
PARALLEL -WIRE TO
40-e0 M. ANTENNA
TO
RECEIVER
SINGLE WIRE
ANTENNA
COAX. LINES TO
IOM. ANT 20M ANT
r
RADIO
55 IL
1
COAXIAL
SWR
INDICATOR
LINE
FROM
XMTR
.001 CERAMICS
Figure 42
L
ANTENNA TUNER AND SWR
INDICATOR
FOR
RANDOM
LENGTH HERTZ ANTENNA
TO
TRANSMITTER THROUGH
HARMONIC FILTER
Figure
41
ALTERNATIVE COAXIAL ANTENNA
COUPLER
This circuit is recommended not only as being most desirable when coaxial lines with
low s.w.r. are being used to feed antenna
systems such as rotatable beams, but when
It also Is desired to feed through open -wire
line to some sort of multi -band antenna for
the lower frequency ranges. The tuned circuit of the antenna coupler is operative only
when using the open -wire feed, and then It
is In operation both for transmit and receive.
in such an application will be found to be adequate, since harmonic attenuation has been
accomplished ahead of the antenna coupler.
However, the circuit will be easier to tune,
although it will not have as great a bandwidth,
if the operating Q is made higher.
An alternative arrangement shown in figure
41 utilizes the antenna coupling tank circuit
only when feeding the coaxial output of the
transmitter to the open -wire feed line (or similar multi -band antenna) of the 40- 80 meter antenna. The coaxial lines to the 10 -meter beam
and to the 20 -meter beam would be fed directly
from the output of the coaxial antenna changeover relay through switch S.
rise to several thousand ohms (near half -wave
resonance) and the reactive component of the
load can rapidly change from positive to negative values, or vice -versa.
It is possible to match a 52 -ohm transmission line to such an antenna at almost any
frequency between 1.8 me and 30 me with the
use of a simple tuner of the type shown in
figure 42. A variable series inductor L, and a
variable shunt capacitor Cl permit circuit
resonance and impedance transformation to
be
established for most antenna lengths.
Switch S1 permits the selection of series
capacitor C for those instances when the
single wire antenna exhibits large values of
positive reactance.
To provide indication for the tuning of the
network, a radio frequency bridge (SWR meter)
is included to indicate the degree of mismatch (standing wave ratio) existing at the
input to the tuner. All adjustments to the
tuner are made with the purpose of reaching
unity standing wave ratio on the coaxial feed
system between the tuner and the transmitter.
A simple antenna tuner for
use with transmitters of
250 watts power or less
is shown in figures 43 through 46. A SWR
bridge circuit is used to indicate tuner resonance. The resistive arm of the bridge consists of ten 10 -ohm, 1 -watt carbon resistors
connected in parallel to form a 1 -ohm resistor
(R1). The other pair of bridge arms are capacitive rather than resistive. The bridge
A
Practical
Antenna Tuner
22 -12
A
Single -Wire Antenna Tuner
One of the simplest and least expensive
antennas for transmission and reception is
the single wire, end -fed Hertz antenna. When
used over a wide range of frequencies, this
type of antenna exhibits a very great range of
input impedance. At the low frequency end of
the spectrum such an antenna may present a
resistive load of less than one ohm to the
transmitter, combined with a large positive or
negative value of reactance. As the frequency
of operation is raised, the resistive load may
detector is
a
simple r -f voltmeter employing a
1N56 crystal diode and a 0 -1 d.c. milliammeter.
A sensitivity control is incorporated to prevent
overloading the meter when power is first
applied to the tuner. Final adjustments are
made with the sensitivity control at its maxi-
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HANDBOOK
Antenna
ter, assuming that the antenna feed line is being operated with a low standing -wave ratio.
However, there are many cases where it is desirable to feed a multi -band antenna from the
output of the harmonic filter, where a tuned
line is being used to feed the antenna, or
where a long wire without a separate feed line
is to be fed from the output of the harmonic
filter. In such cases an antenna coupler is required.
Some harmonic attenuation will be provided
by the antenna coupler, particularly if it is
well shielded. In certain cases when a pi network is being used at the output of the transmitter, the addition of a shielded antenna coupler will provide sufficient harmonic attenuation. But in all normal cases it will be necessary to include a harmonic filter between the
output of the transmitter and the antenna coupler. When an adequate harmonic filter is being used, it will not be necessary in normal
cases to shield the antenna coupler, except
from the standpoint of safety or convenience.
Function of an
Antenna Coupler
The function of the antenna
coupler is, basically, to
transform the impedance of
the antenna system being used to the correct
value of resistive impedance for the harmonic
filter, and hence for the transmitter. Thus the
antenna coupler may be used to resonate the
feeders or the radiating portion of the antenna
system, in addition to its function of imped-
ance transformation.
It is important to remember that there is
nothing that can be done at the antenna coupler which will eliminate standing waves on
the antenna transmission line. Standing waves
are the result of reflection from the antenna,
and the coupler can do nothing about this condition. However, the antenna coupler can resonate the feed line (by introducing a conjugate
impedance) in addition to providing an impedance transformation. Thus, a resistive impedance of the correct value can be presented to
the harmonic filter, as in figure 36, regardless
of any reasonable value of standing -wave ratio
on the antenna transmission line.
All usual types of antenna
couplers fall into two classifications: (1) inductively
coupled resonant systems as exemplified by
those shown in figure 39, and (2) conductively
coupled pi- network systems such as shown in
figure 40. The inductively -coupled system is
much more commonly used, since it is convenient for feeding a balanced line from the coTypes of
Antenna Couplers
axial output of the usual harmonic filter. The
pi- network system is most useful for feeding
a length of wire from the output of a transmitter.
Couplers
COAX TO
RECEIVER
TRANSMITTER
HARMONIC
FILTER
451
SINGLE WIRE
ANTENNA
COAX ANT.
CHANGEOVER
RELAY
Figure 40
PI- NETWORK
ANTENNA COUPLER
An arrangement such as illustrated
above is convenient for feeding an
end -fed Hertz antenna, or a random
length of wire for portable or emergency operation, from the nominal
value of impedance of the harmonic
filter.
Several general methods for using the inductively- coupled resonant type of antenna coupler are illustrated in figure 39. The coupling
between the link coil L and the main tuned circuit need not be variable; in fact it is preferable that the correct link size and placement
be determined for the tank coil which will be
used for each band, and then that the link be
made a portion of the plug -in coil. Capacitor
C then can be adjusted to a pre- determined
value for each band such that it will resonate
with the link coil for that band. The reactance
of the link coil (and hence the reactance of the
capacitor setting which will resonate the coil)
should be about 3 or 4 times the impedance of
the transmission line between the antenna coupler and the harmonic filter, so that the link
coupling circuit will have an operating Q of
3 or 4. The use of capacitor C to resonate with
the inductance of the link coil L will make it
easier to provide a low standing -wave ratio
to the output of the harmonic filter, simply by
adjustment of the antenna- coupler tank circuit
to resonance. If this capacitor is not included,
the system still will operate satisfactorily, but
the tank circuit will have to be detuned slightly from resonance so as to cancel the inductive reactance of the coupling link and thus
provide a resistive load to the output of the
harmonic filter. Variations in the loading of
the final amplifier should be made by the coupling adjustment at the final amplifier, not at
the antenna coupler.
The pi- network type of antenna coupler, as
shown in figure 40 is useful for certain applications, but is primarily useful in feeding a
single -wire antenna from a low- impedance
transmission line. In such an application the
operating Q of the pi network may be somewhat
lower than that of a pi network in the plate circuit of the final amplifier of a transmitter, as
shown in figure 38. An operating Q of 3 or 4
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450
THE
Antennas and Antenna Matching
RADIO
COAX. TO
RECEIVER
TRANSMITTER
U
HARMONIC
FILTER
LJ
¡
COAX ANT.
CNANGEOVEq
RELAY
PARALLEL-WIRE
LINE TO ANTENNA
L
SINGLE -WIRE
HERTZ ANTENNA
OR FEEDER
CEPP
FEEDERS
O
SINGLE-WIRE
COAX. LINE
ANTENNA OR
FEEDLINE
TO ANTENNA
O
©
Figure
39
ALTERNATIVE ANTENNA -COUPLER CIRCUITS
Plug -in coils, one or two variable capacitors of the split-stator variety, and a system of
switches or plugs and jacks may be used in the antenna coupler to accomplish the feeding of
different types of antennas and antenna transmission lines from the coaxial input line from the
transmitter or from the antenna changeover relay. Link L should be resonated with capacitor
C at the operating frequency of the transmitter so that the harmonic filter will operate into o
resistive load impedance of the correct nominal value.
ended output stage down to the 50 -ohm impedance of the usual harmonic filter and its sub -
sequent load.
In a pi network of this type the harmonic
attenuation of the section will be adequate
when the correct value of C, and L are being
used and when the r e s on a n t dip in C, is
sharp. If the dip in C, is broad, or if the plate
current persists in being too high with C2 at
maximum setting, it means that a greater value
of capacitance is required at C2, assuming that
the values of C, and L are correct.
22 -11
Antenna Couplers
As stated in the previous section, an antenna coupler is not required when the impedance
of the antenna transmission line is the same
as the nominal impedance of the harmonic fil-
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HANDBOOK
Pi- Network Coupling Systems
449
COAX. TO
RECEIVER
i
SHIELO
HARMONIC
oAK---- ATTENUATING
C
1
COAX
1
ANT.
CHAANGEOVER
1
- - -I
1
ANTENNA
FEEDLINE OR
CO ANTENNA
COUPLER
TO
Figure 37
TUNED -LINK OUTPUT CIRCUIT
Capacitor C should be adjusted so os to tune out the inductive reactance of the coupling link,
L. Loading of the amplifier then is varied by physically varying the coupling between the plate
tank of the final amplifier and the antenna coupling link,
The pi- network coupling system
offers two advantages: (1) a mechanical coupling variation is
not required to vary the loading of the final amplifier, and (2) the pi network (if used with an
operating Q of about 15) offers within itself a
harmonic attenuation of 40 db or more, in addition to the harmonic attenuation provided by
the additional harmonic attenuating filter. Some
Pi- Network
Coupling
commercial equipments (such as the Collins
amateur transmitters) incorporate an L network
in addition to the pi network, for accomplishing the impedance transformation in two steps
and to provide additional harmonic attenuation.
Tuning the
Pi- Section Coupler
Tuning of a pi- network
coupling circuit such as
illustrated in figure 38 is
accomplished in the following manner: First
remove the connection between the output
of
the amplifier and the harmonic filter (load).
Tune C2 to a capacitance which is large for
the band in use, adding suitable additional ca-
pacitance by switch S if operation is to be on
one of the lower frequency bands. Apply reduced plate voltage to the stage and dip to
resonance with C,. It may be necessary to vary
the inductance in coil L, but in any event resonance should be reached with a setting of C,
which is approximately correct for the desired
value of operating Q of the pi network.
Next, couple the load to t h e amplifier
(through the harmonic filter), apply reduced
plate voltage again and dip to resonance with
C,. If the plate current dip with load is too low
(taking into consideration the reduced plate
voltage), decrease the capacitance of C2 and
again dip to resonance, repeating the procedure until the correct value of plate current is
obtained with full plate voltage on the stage.
There should be a relatively small change required in the setting of C, (from the original
setting of C, without load) if the operating Q
of the network is correct and if a large value
of impedance transformation is being employed-as would be the case when transforming from the plate impedance of a singleCOAX TO
RECEIVER
HARMONIC
_ATTENUATING
FILTER
=
r
COAXIAL
ANTENNA
,CHANGEOVER
RELAY
TO FCCDLINE
-MOR
ANTENNA
COUPLER
Figure 38
PI- NETWORK ANTENNA COUPLER
The design of pi-network output circuits is discussed
end shunting capacitors selected by switch S are for in Chapter Thirteen. The additional output use on the lower frequency ranges. Inductor L may be selected by a tap switch, it may be continuously
variable, or plug -in inductors
may be used.
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448
THE
Antennas and Antenna Matching
AT TRANSMITTER
I
EXCITER
PORTION
rI NAL
AMPLIFIE
COUPLING
RDJUSTMENT
HARMONIC
ATTENUATING
SYSTEM
ANTENNA
COUPLER
RADIO
AT ANTENNA
IMPEDANCE
TRANSMISSION
LIN[
MATCHING
TANTENNA
RADIATING
SYSTEM
Figure 36
ANTENNA COUPLING SYSTEM
The antenna coupling system illustrated above is for use when the antenna transmission line
does not have the same characteristic impedance as the TVI filter, and when the standing -wave
ratio on the antenna transmission line may or may not be low.
within or adjacent to the antenna. The feed
line coming down from the an t e n n a system
should have a characteristic impedance equal
to the nominal impedance of the harmonic filter, and the impedance matching at the antenna should be such that the standing -wave ratio
on the antenna feed line is less than 2 to 1
over the range of frequency to be fed to the
antenna. Such an arrangement may be used
with open -wire line, ribbon or tubular line, or
with coaxial cable. The use of coaxial cable
is to be recommended, but in any event the
impedance of the antenna transmission line
should be the same as the nominal impedance
of the harmonic filter. The arrangement of figure 35 is more or less standard for commercially manufactured equipment for amateur and
commercial use in the h -f and v -h -f range.
The arrangement of figure 36 merely adds
an antenna coupler between the output of the
harmonic attenuating filter and the antenna
transmission line. The antenna coupler will
have some harmonic- attenuating action, but its
main function is to transform the impedance
at the station end of the antenna transmission
line to the nominal value of the harmonic filter.
Hence the arrangement of figure 36 is more
general than the figure 35 system, since the
inclusion of the antenna coupler allows the
system to feed an antenna transmission line
of any reasonable impedance value, and also
without regard to the standing -wave ratio
which might exist on the antenna transmission
line. Antenna couplers are discussed in a following section.
It will be noticed by reference to both figure 35 and
figure 36 that a box labeled
Coupling Adjustment is included in the block
Output Coupling
Adjustment
diagram. Such an element is necessary in the
complete system to afford an adjustment in the
value of load impedance presented to the tubes
in the final amplifier stage of the transmitter.
The impedance at the input terminal of the
harmonic filter is established by the antenna,
through its matching system and the antenna
coupler, if used. In any event the impedance
at the input terminal of the harmonic filter
should be very close to the nominal impedance
of the filter. Then the Coupling Adjustment
provides means for transforming this impedance value to the correct operating value of
load impedance which should be presented to
the final amplifier stage.
There are two common ways for accomplishing the antenna coupling adjustment, as illustrated in figures 37 and 38. Figure 37 shows
the variable -link arrangement most commonly
used in home -constructed equipment, while the
pi- netowrk coup ling arrangement commonly
used in commercial equipment is illustrated in
figure 38. Either method may be used, and each
has its advantages.
Variable-Link
Coupling
variable -link method illustrated in figure 37 has the
advantage that standard manufactured components m a y be used with no
changes. However, for greatest bandwidth of
operation of the coupling circuit, the reactance
of the link coil, L, and the reactance of the
link tuning capacitor, C, should both be beThe
tween 3 and 4 times the nominal load impedance of the harmonic filter. This is to say that
the inductive reactance of the coupling link L
should be tuned out or resonated by capacitor
C, and the operating Q of the L -C link circuit
should be between 3 and 4. If the link coil is
not variable with respect to the tank coil of
the final amplifier, capacitor C may be used
as a loading control; however, this system is
not recommended since its use will require
adjustment of C whenever a frequency change
is made at the transmitter. If L and C are made
resonant at the center of a band, with a link
circuit Q of 3 to 4, and coupling adjustment is
made by physical adjustment of L with respect
to the final amplifier tank coil, it usually will
be possible to operate over an entire amateur
band without change in the coupling system.
Capacitor C normally may have a low voltage
rating, even with a high power transmitter, due
to the low Q and low impedance of the coupling
circuit.
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Antenna Coupling Systems
HANDBOOK
When insulators are subject to very high r -f
voltages, they should be cleaned occasionally
if in the vicinity of sea water or smoke. Salt
scum and soot are not readily dislodged by
rain, and when the coating becomes heavy
enough, the efficiency of the insulators is
greatly impaired.
If a very pretentious installation is to be
made, it is wise to check up on both underwriter's rules and local ordinances which
might be applicable. If you live anywhere near
an airport, and are contemplating a tall pole,
it is best to investigate possible regulations
and ordinances pertaining to towers in the district, before starting construction.
Coupling to the
Antenna System
22 -10
When coupling an antenna feed system to a
transmitter the most important considerations
are as follows: (1) means should be provided
for varying the load on the amplifier; (2) the
two tubes in a push -pull amplifier should be
equally loaded; (3) the load presented to the
final amplifier should be resistive (non -reactive) in character; and (4) means should be
provided to reduce harmonic coupling between
the final amplifier plate tank circuit and the
antenna or antenna transmission line to an extremely low value.
The problem of coupling the
power output of a high -frequency or v -h -f transmitter
to the radiating portion of the antenna system
has been materially complicated by the virtual
necessity for eliminating interference to TV reception. However, the TVI- elimination portion
of the problem may always be accomplished
by adequate shielding of the transmitter, by
filtering of the control and power leads which
enter the transmitter enclosure, and by the inclusion of a harmonic -attenuating filter beThe TransmitterLoading Problem
1NIELD
rE%CITER
PORTION
FINAL
AMPLIFIER
tween the output of the transmitter and the antenna system.
Although TVI may be eliminated through inclusion of a filter between the output of a
shielded transmitter and the antenna system,
the fact that such a filter must be included in
the link between transmitter and antenna makes
it necessary that the transmitter -loading problem be re- evaluated in terms of the necessity
for inclusion of such a filter.
Harmonic- attenuating filters must be operated at an impedance level which is close to
their design value; therefore they must operate
into a resistive termination substantially equal
to the characteristic impedance of the filter.
If such filters are operated into an impedance
which is not resistive and approximately equal
to their characteristic impedance: (1) the capacitors used in the filter sections will be
subjected to high peak voltages and may be
damaged, (2) the harmonic- attenuating properties of the filter will be decreased, and (3) the
impedance at the input end of the filter will
be different from that seen by the filter at the
load end (except in the case of the half-wave
type of filter). It is therefore important that
the filter be included in the transmitter-to-an tenna circuit at a point where the impedance
is close to the nominal value of the filter, and
at a point where this impedance is likely to
remain fairly constant with variations in frequency.
There are two basic
arrangements which
include all the provisions required in the
transmitter-to- antenna coupling system, and
which permit the harmonic -attenuating filter to
be placed at a position in the coupling system
where it can be operated at an impedance
level close to its nominal value. These arrangements are illustrated in block -diagram
form in figures 35 and 36.
The arrangement of figure 35 is recommended for use with a single -band antenna system,
such as a dipole or a rotatable array, wherein
an impedance matching system is included
Block Diagrams of
Transmitter -to- Antenna
Coupling Systems
AT TRANSMITTER
COUPLING
Ap1U5TMENT
_ HARMONIC
TTENUATI
SYSTEM
447
AT ANTENNA
I
TRANSMISSION
LINE
MPE DANCED RADIATING
MATCHING
T
ANTENNAn SYSTEM
Figure 35
ANTENNA COUPLING SYSTEM
The harmonic suppressing antenna coupling system illustrated above is for use when the antenna transmission line has a low standing -wave ratio, and when the characteristic impedance of
the antenna transmission line is the same as the nominal impedance of the low -pass harmonicattenuating filter.
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4 4 6
Antennas and Antenna
M
waiting for it to show excessive wear or deterioration.
It is an excellent idea to tie both ends of
the halyard line together in the manner of a
flag -pole line. Then the antenna is tied onto
the place where the two ends of the halyard
are joined. This procedure of making the halyard into a loop prevents losing the top end
of the halyard should the antenna break near
the end, and it also prevents losing the halyard completely should the end of the halyard
carelessly be allowed to go free and be pulled
through the pulley at the top of the mast by
the antenna load. A somewhat longer piece
of line is required but the insurance is well
worth the cost of the additional length of rope.
Often a tall tree can be called upon to support one end of an antenna, but one should not attempt to
attach anything to the top, as the swaying of
the top of the tree during a heavy wind will
complicate matters.
If a tree is utilized for support, provision
should be made for keeping the antenna taut
without submitting it to the possibility of being severed during a heavy wind. This can be
done by the simple expedient of using a pulley and halyard, with weights attached to the
lower end of the halyard to keep the antenna
taut. Only enough weight to avoid excessive
sag in the antenna should be tied to the halyard, as the continual swaying of the tree submits the pulley and halyard to considerable
wear.
Galvanized iron pipe, or steel -tube conduit,
is often used as a vertical radiator, and is
quite satisfactory for the purpose. However,
when used for supporting antennas, it should
be remembered that the grounded supporting
poles will distort the field pattern of a vertically polarized antenna unless spaced some
distance from the radiating portion.
Trees as
Supports
The life of a wood mast or pole can
be increased several hundred per
cent by protecting it from the elements with a
coat or two of paint. And, of course, the appearance is greatly enhanced. The wood should
first be given a primer coat of flat white outside house paint, which can be thinned down
a bit to advantage with second -grade linseed
oil. For the second coat, which should not be
applied until the first is thoroughly dry, aluminum paint is not only the best from a preservative standpoint, but looks very well. This type
of paint, when purchased in quantities, is considerably cheaper than might be gathered from
the price asked for quarter-pint cans.
Portions of posts or poles below the surface
of the soil can be protected from termites and
moisture by painting with creosote. While not
Painting
THE
atching
RADIO
so strong initially, redwood will deteriorate
much more slowly when buried than will the
white woods, such as pine.
The
Antenna Wire
antenna
or array
itself
presents no especial problem.
A few considerations should be borne in mind,
however. For instance, soft -drawn copper
should not be used, as even a short span will
stretch several per cent after whipping around
in the wind a few weeks, thus affecting the
resonant frequency. Enameled -copper wire,
as ordinarily available at radio stores, is usally soft drawn, but by tying one end to some
object such as a telephone pole and the other
to the frame of an auto, a few husky tugs can
be given and the wire, after stretching a bit,
is equivalent to hard drawn.
Where a long span of wire is required, or
where heavy insulators in the center of the
span result in considerable tension, copper clad steel wire is somewhat better than harddrawn copper. It is a bit more expensive,
though the cost is far from prohibitive. The
use of such wire, in conjunction with strain
insulators, is advisable, where the antenna
would endanger persons or property should it
break.
For transmission l i n e s and tuning stubs
steel -core or hard -drawn wire will prove awkward to handle, and soft -drawn copper should,
therefore, be used. If the line is Ion g, the
strain can be eased by supporting it at several
points.
More important from an electrical standpoint
than the actual size of wire used is the soldering of joints, especially at current loops
in an antenna of low radiation resistance. In
fact, it is good practice to solder all joints,
thus insuring quiet operation when the antenna is used for receiving.
question that often arises is
that of insulation. It depends, of
course, upon the r-f voltage at the point at
which the insulator is placed. The r -f voltage,
in turn, depends upon the distance from a current node, and the radiation resistance of the
antenna. Radiators having low radiation resistance have very high voltage at the voltage
loops; consequently, better than usual insulation is advisable at those points.
Open -wire lines operated as non -resonant
lines have little voltage across them; hence
the most inexpensive ceramic types are sufficiently good electrically. With tuned lines, the
Insulation
A
voltage depends upon the amplitude of the
standing waves. If they are very great, the
voltage will reach high values at the voltage
loops, and the best spacers available are none
too good. At the current loops the voltage is
quite low, and almost anything will suffice.
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Antenna
HANDBOOK
Construction
445
2X4
Figure 34
SIMPLE WOOD MASTS
Shown at (A) is the method of assembly, and at (B) is the completed
structure, of the conventional "Aframe" antenna most. At (C) is
shown a structure which is heavier
but more stable than the A -frame
for heights above about 40 feet.
TWO
SAWHORSES
/-\\
Iv-S
2X2
CROSSPIECES
GROUND LEVEL
i
I
CONCRETE éd
'9JJéi
s'
©
if a gin pole about 20 feet high is installed
about 30 or 40 feet to the rear of the direction
in which the antenna is to be raised. A line
from a pulley on the top of the gin pole is then
run to the top of the pole to be raised. The
gin pole comes into play when the center of
the mast has been raised 10 to 20 feet above
the ground and an additional elevated pull is
required to keep the top of the mast coming
up as the center is raised further above
ground.
Steel tubing masts of the
telescoping variety are widely available at a moderate price for use in supporting television antenna arrays. These masts
usually consist of several 10-foot lengths of
electrical metal tubing (EMT) of sizes such
that the sections will telescope. The 30 -foot
and 40-foot lengths are well suited as masts
for supporting antennas and arrays of the
types used on the amateur bands. The masts
are constructed in such a manner that the bottom 10-foot length may be guyed permanently
before the other sections are raised. Then the
upper sections may be extended, beginning
with the top -mast section, until the mast is at
full length(provided a strong wind is not blowing) following which all the guys may be anchored. It is important that there be no load
on the top of the mast when the "vertical"
raising method is to be employed.
Using TV Masts
Guy Wires
Guy wires should never be pulled
taut; a small amount of slack is
desirable. Galvanized wire, somewhat heavier
than seems sufficient for the job, should be
used. The heavier wire is a little harder to
handle, but costs only a little more and takes
longer to rust through. Care should be taken
to make sure that no kinks exist when the pole
or tower is ready for erection, as the wire will
be greatly weakened at such points if a kink
is pulled tight, even if it is later straightened.
If "dead men" are used for the guy wire
terminations, the wire or rod reaching from the
dead men to the surface should be of non -rusting material, such as brass, or given a heavy
coating of asphalt or other protective substance to prevent destructive action by the
damp soil. Galvanized iron wire will last only
a short time when buried in moist soil.
Only strain -type (compression) insulators
should be used for guy wires. Regular ones
might be sufficiently strong for the job, but it
is not worth taking chances, and egg -type
strain halyard insulators are no more ex-
pensive.
Only a brass or bronze p u l l e y should be
used for the halyard, as a high pole with a
rusted pulley is truly a sad affair. The bearing of the pulley should be given a few drops
of heavy machine oil before the pole or tower
is raised. The halyard itself should be of good
material, preferably water-proofed. Hemp rope
of good quality is better than window sash
cord from several standpoints, and is less expensive. Soaking it thoroughly in engine oil of
medium viscosity, and then wiping it off with
a rag, will not only extend its life but minimize shrinkage in wet weather. Because of the
difficulty of replacing a broken halyard it is
a good idea to replace it periodically, without
www.americanradiohistory.com
444
ance of 123 ohms. Z, is one -quarter wavelength
long at the mid-frequency and has an impedance of 224 ohms.
is the balanced line to
be matched(in this case 300 ohms) and may be
any length.
4
Other system parameters for different output
and input impedances may be calculated from
the following:
Transformation ratio (r) for each section is:
r=
Z
N
Zrn
Where N
is the number of sections. In the
above case,
Z5
3
Z
r
Impedance between sections, as
times the preceding section. Z=_, = r
= r X
Z,_,
Z,_
X
is
r
Z and
Mid -frequency (m):
m=
F, + F,
For 40 -20 -10 meters =
+30
7
-
18.5 Mc.
2
and one -quarter wavelength = 12 feet.
For 20 -10-6 meters
--
14 + 54
-
34 Mc.
2
and one -quarter wavelength = 5.5 feet.
The impedances of the sections are:
Z, =V Z,
Z,
=
Z4
-7-
Zo=
X
V Z:-,
Z,_,
X
RADIO
been concerned primarily with the electrical
characteristics and considerations of antennas. Some of the physical aspects and mechanical problems incident to the actual erection of antennas and arrays will be discussed
in the following section.
Up to 60 feet, there is little point in using
mast -type antenna supports unless guy wires
either must be eliminated or kept to a minimum. while a little more difficult to erect, because of their floppy nature, fabricated wood
poles of the type to be described will be just
as satisfactory as more rigid types, provided
many guy wires are used.
Rather expensive when purchased through
the regular channels, 40- and 50 -foot telephone poles sometimes can be obtained quite
reasonably. In the latter case, they are hard
to beat, inasmuch as they require no guying
if set in the ground six feet (standard depth),
and the resultant pull in any lateral direction
is not in excess of a hundred pounds or so.
For heights of 80 to 100 feet, either three sided or four -sided lattice type masts are most
practicable. They can be made self- supporting, but a few guys will enable one to use a
smaller cross section without danger from
high winds. The torque exerted on the base
of a high self- supporting mast is terrific dur-
ing a strong wind.
2
Z3-4
\7Z74.727
XZ,
Generally, the larger number of taper sections the greater will be the bandwidth of the
system.
22 -9
THE
Antennas and Antenna Matching
Antenna
Construction
The foregoing portion of this chapter has
The "A- Frame" Figures 34A and 34B show
the standard method of conMost
struction of the A -frame
type of mast. This type of mast is quite frequently used since there is only a moderate
amount of work involved in the construction
of the assembly and since the material cost is
relatively small. The three pieces of selected
2 by 2 are first set up on three sawhorses or
boxes and the holes drilled for the three 1/4inch bolts through the center of the assembly.
Then the base legs are spread out to about 6
feet and the bottom braces installed. Then the
upper b r aces and the cross pieces are installed and the assembly given several coats
of good -quality paint as a protection against
weathering.
Figure 34C shows another common type of
mast which is made up of sections of 2 by 4
placed end -to -end with stiffening sections of
1 by
6 bolted to the edge of the 2 by 4 sections. Both types of masts will require a set
of top guys and another set of guys about one third of the way down from the top. Two guys
spaced about 90 to 100 degrees and pulling
against the load of the antenna will normally
be adequate for the top guys. Three guys are
usually used at the lower level, with one directly behind the load of the antenna and two
more spaced 120 degrees from the rear guy.
The raising of the mast is made much easier
www.americanradiohistory.com
Matching Systems
HANDBOOK
Ì`
-
L FEET
=
_
Center to
Center
Spacing
in Inches
234
Zo=
FIMCI
Impedance
in Ohms
for %z"
Diameters
for I/4 "
Diameters
170
188
207
225
248
1.25
1.5
1.75
Z2
Impedance
in Ohms
1
Q MATCHING SECTION
TUBING
L
46B
F Mc
443
2
250
277
298
318
335
PARALLEL TUBING SURGE IMPEDANCE FOR
So
MATCHING SECTIONS
z
The Collins
Transmission Line
UNTUNED LINE
ANY LENGTH
ally can be obtained by either designing or
adjusting the matching section for a dipole to
a
fo r transmitters a r e numerous; however this output system becomes awkward when it is desired to feed an antenna system utilizing a
balanced i n p u t. For some time the Collins
Radio Co. has been experimenting with a balun and tapered line system for matching a coaxial output transmitter to an open -wire balanced transmission line. Considerable success
has been obtained and matching systems good
over a frequency range as great as four to one
have been developed. Illustrated in figure 33
is one type of matching system which is proving satisfactory over this range. Z, is the
transmitter end of the system and may be any
length of 52 -ohm coaxial cable. Z2 is one quarter wavelength long at the mid-frequency
of the range to be covered and is made of 75
ohm coaxial cable. ZA is a quarter- wavelength
shorted section of cable at the mid -frequency.
ZD (ZA and Z2) forms a 200-ohm quarter -wave
section. The ZA section is formed of a conductor of the same diameter as Z2. The difference in length between ZA and Z2 is accounted
for by the fact that Z2 is a coaxial conductor
with a solid dielectric, whereas the dielectric
for ZD is air. Z2 is one -quarter wavelength
long at the mid -frequency and has an impedMatching System
Figure 32
HALF -WAVE RADIATOR FED
BY "Q BARS"
The Q matching section is simply o quarter wave transformer whose impedance is equal
to the geometric mean between the impedance at the center of the antenna and the
impedance of the transmission line to be
used to feed the bottom of the transformer.
The transformer may be made up of parallel
tubing, a four -wire line, or any other type
of transmission line which has the correct
value of impedance.
have
The advantage of unbalanced output networks
surge impedance that is the geometric
mean between the line impedance and 72 ohms,
the latter being the theoretical radiation re-
sistance of a half -wave doublet either infinitely high or a half wave above a perfect
ground.
Though the radiation resistance may depart
somewhat from 72 ohms under actual condi-
tions, satisfactory results will be obtained
with this assumed value,so long as the dipole
radiator is more than a quarter wave above
effective earth, and reasonably in the clear.
The small degree of standing waves introduced by a slight mismatch will not increase
the line losses appreciably, and any small
amount of reactance present can be tuned out
at the transmitter termination with no bad effects. If the reactance is objectionable, it may
be minimized by making the untuned line an
integral number of quarter waves long.
A Q- matched system can be adjusted precisely, if desired, by constructing a matching
section to the calculated dimensions with provision for varying the spacing of the Q section conductors slightly, after the untuned
line has been checked for standing waves.
Zo
ZJ
Z4
If__,_,,./____n____
Zz
ZI
----
2J=123R
v
ZA
ZS
I
Z= 22411
Z5=30011
(ANY LENGTH)
}
INNER L OUTER CONDUCTORS
SHORTED AT EACH END
Figure 33
COLLINS TRANSMISSION LINE MATCHING
SYSTEM
wide-band system for matching a 52 -ohm
coaxial line to a balanced 300 -ohm line over
a 4:7 wide frequency range.
A
www.americanradiohistory.com
442
The open stub should be resonated in the
same manner as the shorted stub before attaching the transmission line; however, in
this case, it is necessary to prune the stub
to resonance, as there is no shorting bar.
Sometimes it is handy to have a stub hang
from the radiator to a point that can be reached
from the ground, in order to facilitate adjustment of the position of the transmission -line
attachment. For this reason, a quarter -wave
stub is sometimes made three -quarters wavelength long at the higher frequencies, in order
to bring the bottom nearer the ground. Operation with any odd number of quarter waves is
the same as for a quarter -wave stub.
Any number of half waves can be added to
either a quarter-wave stub or a half -wave stub
without disturbing the operation, though losses
and frequency sensitivity will be lowest if
the shortest usable stub is employed. See figure 31.
Stub Length
(Electrical)
%.etc.
1/4- % -1 t
wavelengths
Y2 -1 -1 i5.2 -etc .
wavelength s
Linear
THE
Antennas and Antenna Matching
R -F
Transformers
Current -Fed
Radiator
Voltage -Fed
Radiator
Open
Stub
Shorted
Stub
Shorted
Stub
Open
Stub
resonant quarter -wave line
the unusual property of
acting much as a transformer.
Let us take, for example, a section consisting
of no. 12 wire spaced 6 inches, which happens
to have a surge impedance of 600 ohms. Let
the far end be terminated with a pure resistance, and let the near end be fed with radio frequency energy at the frequency for which
the line is a quarter wavelength long. If an
impedance measuring set is used to measure
the impedance at the near end while the impedance at the far end is varied, an interesting relationship between the 600 -ohm characteristic surge impedance of this particular
quarter -wave matching line, and the impedance at the ends will be discovered.
When the impedance at the far end of the
line is the same as the characteristic surge
impedance of the line itself (600 ohms), the
impedance measured at the near end of the
quarter -wave line will also be found to be
A
has
600 ohms.
Under these conditions, the line would not
have any standing waves on it, since it is
terminated in its characteristic impedance.
Now, let the resistance at the far end of the
line be doubled, or changed to 1200 ohms.
The impedance measured at the near end of
the line will be found to have been cut in
RADIO
half, to 300 ohms. If the resistance at the far
end is made half the original value of 600
ohms or 300 ohms, the impedance at the near
end doubles the original value of 600 ohms,
and becomes 1200 ohms. As one resistance
goes up, the other goes down proportionately.
It will always be found that the characteristic surge impedance of the quarter -wave
matching line is the geometric mean between
the impedance at both ends. This relationship
is shown by the following formula:
ZMS
=
ZA ZL
where
= Impedance of matching
= Antenna resistance.
= Line impedance.
ZMS
ZA
ZL
section.
The impedance inverting char-
Quarter -Wave
Matching
Transformers
acteristic
of a quarter -wave
section of transmission line is
widely used by making such a
section of line act as a quarter-wave transformer. The Johnson Q feed system is a widely known application of the quarter-wave transformer to the feeding of a dipole antenna and
array consisting of two dipoles. However, the
quarter -wave transformer may be used in a
wide number of applications wherever a transformer is required to match two impedances
whose geometric mean is somewhere between
perhaps 25 and 750 ohms when transmission
line sections can be used. Paralleled coaxial
lines may be used to obtain the lowest impedance mentioned, and open -wire lines composed of small conductors spaced a moderate
distance may be used to obtain the higher impedance. A short list of impedances, which
may be matched by quarter -wave sections of
transmission line having specified impedances, is given below.
Load or Ant.
Impedance
1
20
30
50
75
100
300
77
95
110
150
173
480
600
98
120
139
190
220
110
134
155
212
r
Feed -Line
Impedance
Quarter Wave
Transformer
Impedance
245
The standard form of Johnson n feed to a doublet is shown
in figure 32. An impedance
match is obtained by utilizing a matching section, the surge impedance of which is the geometric mean between the transmission line
surge impedance and the radiation resistance
of the radiator. A sufficiently good match usuJohnson -Q
Feed System
www.americanradiohistory.com
HANDBOOK
Matching Systems
441
section. If they are not used, the T- section
will detune the dipole when the T- section is
attached to it. The two capacitors may be
ganged together, and once adjusted for minimum detuning action, they may be locked. A
suitable housing should be devised to protect
these capacitors from the weather. Additional
information on the adjustment of the T -match
is given in the chapter covering rotary beam
antennas.
SHORTING BAR
ANTENNA
unbalanced version of
the T -match may be used
to feed a dipole from an unbalanced coaxial
line. Such a device is called a Gamma Match,
and is illustrated in figure 30.
The length of the Gamma rod and the spacing of it from the dipole determine the impedance level at the transmission line end of the
rod. The series capacitor is used to tune out
the reactance introduced into the system by
the Gamma rod. The adjustment of the Gamma
Match is discussed in the chapter covering
rotary beam antennas.
The "Gamma" Match
An
connecting a resonant
section of transmission line
(called a matching stub) to either a voltage or
current loop and attaching parallel -wire non resonant feeders to the resonant stub at a
suitable voltage (impedance) point, standing
waves on the line may be virtually eliminated.
The stub is made to serve as an auto- transformer. Stubs are particularly adapted to
matching an open line to certain directional
arrays, as will be described later.
Matching Stubs
SHORTING BAR
NON-RESONANT
FEEDERS
ANTENNA
FEEDER TAPS NEAR
END OF STUB
NON- RESONANT
FEEDERS
STUB
When the
When a stub is used to current feed a radiator, the stub should
either be left open at the bottom end instead
of shorted, or else made a half wave long.
Current Feed
STUB
By
stub attaches to the
antenna at a voltage loop, the
stub should be a quarter wavelength long
electrically, and be shorted at the bottom end.
The stub can be resonated by sliding the
shorting bar up and down before the non -resonant feeders are attached to the stub, the antenna being shock -excited from a separate
radiator during the process. Slight errors in
the length of the radiator can be compensated
for by adjustment of the stub if both sides of
the stub are connected to the radiator in a
symmetrical manner. Where only one side of
the stub connects to the radiating system, as
in the Zepp and in certain antenna arrays, the
radiator length must be exactly right in order
to prevent excessive unbalance in the untuned
line.
A dial lamp may be placed in the center of
the shorting stub to act as an r -f indicator.
Voltage Feed
11
OPEN
t7
STUB
NON- RESONANTSHORTING BAR
FEEDER
Figure 31
MATCHING -STUB APPLICATIONS
An end -fed half-wave antenna with a quarterwave shorted stub is shown at (A). (B) shows
the use of a half-wave shorted stub to feed
a relatively low impedance point such as the
center of the driven element of a parasitic
array, or the center of a half-wave dipole.
The use of an open -ended quarter -wave stub
to feed a low impedance is illustrated at
(C). (D) shows the conventional use of a
shorted quarter -wave stub to voltage feed
two half-wave antennas with a 180. phase
difference.
www.americanradiohistory.com
440
THE
Antennas and Antenna Matching
RADIO
DRIVEN ELEMENT
MOVEABLE CLAMP
GAMMA ROD
RESONATING CONDENSER
-.:7°-LE.D-
I7
EL.T-T0. 0.
50 -70 OHM COAXIAL FEED LINE
I
TD
irrN-LE.D
IE1
l000HU.
ANY LENGTN
Figure 30
SOIVEN ELEMENT
GAMMA MATCH
FOR CONNECTING
AN UNBALANCED COAXIAL LINE TO A
THE
BALANCED DRIVEN ELEMENT
/ /
SOD-.00
OHM ECLDC03
l00-.00
ON
Figure 29
FOLDED -ELEMENT MATCHING SYSTEMS
Drawing (A) above shows a half-wave made
up to two parallel wires. If one of the wires
is broken as in (B) and the feeder connected,
the feed -point impedance is multiplied by
four; such an antenna is commonly called a
"folded doublet." The feed-point impedance
for a simple half -wave doublet fed in this
manner is approximately 300 ohms, depending upon antenna height. Drawing (C) shows
how the feed -point impedance can be multiplied by a factor greater than four by making
the half of the element that is broken smaller
in diameter than the unbroken half. An extension of the principles of (B) and (C) is
the arrangement shown at (D) where the section into which the feeders are connected is
considerably shorter than the driven element.
This system is most convenient when the
driven element is too long (such as for o
28 -Mc. or 14 -Mc. array) for a convenient
mechanical arrangement of the system shown
at (C).
wire of such a radiator, as shown in figure 29,
the effective feed-point resistance of the antenna or array will be increased by a factor
of N2 where N is equal to the number of conductors, all in parallel, of the same diameter
in the array. Thus if there are two conductors
of the same diameter in the driven element or
the antenna the feed -point resistance will be
multiplied by 22 or 4. If the antenna has a radiation resistance of 75 ohms its feed -point
resistance will be 300 ohms, this is the case
of the conventional folded- dipole as shown in
figure 29B.
If three wires are used in the driven radiator the feed -point resistance is increased by
a factor of 9; if four wires are used the impedance is increased by a factor of 16, and
so on. In certain cases when feeding a parasitic array it is desirable to have an impedance step up different from the value of 4:1
obtained with two elements of the same diameter and 9:1 with three elements of the same
diameter. Intermediate values of impedance
step up may be obtained by using two elements
of different diameter for the complete driven
element as shown in figure 29C. If the conductor that is broken for the feeder is of smaller diameter than the other conductor of the
radiator, the impedance step up will be greater
than 4:1. On the other hand if the larger of the
two elements is broken for the feeder the impedance step up will be less than 4:1.
"T"
method of matching a balanced low- impedance transmission line to the driven element of a parasitic array is the T match illustrated in figure
29D. This method is an adaptation of the
multi -wire doublet principle which is more
practicable for lower-frequency parasitic axrays such as those for use on the 14 -Mc. and
28 -Mc. bands. In the system a section of tubing of approximately one-half the diameter of
the driven element is spaced about four inches below the driven element by means of
clamps which hold the T- section mechanically
and which make electrical connection to the
driven element. The length of the T- section
is normally between 15 and 30 inches each
side of the center of the dipole for transmission lines of 300 to 600 ohms impedance, assuming 28 -Mc. operation. In series with each
leg of the T- section and the transmission line
is a series resonating capacitor. These two
capacitors tune out the reactance of the TThe
Match
A
Matching Systems
HANDBOOK
-I
D
MATCHING SECTION
439
transmitter end of the feed line which will
change the magnitude of the standing waves
on the antenna transmission line.
The delta type matched -impedance antenna system is
Delta- Marched
Antenna System
shown in figure 28. The im-
NON-RESONANT
LINE
Figure 28
THE
DELTA- MATCHED DIPOLE
ANTENNA
The dimensions for the portions of the an.
tenno ore given in the text.
between three types of transmission line: (1)
Ribbon or tubular molded 300 -ohm line is
widely used up to moderate power levels (the
"transmitting" type is useable up to the kilowatt level). (2) Open -wire 400 to 600 ohm line
is most commonly used when the antenna is
some distance from the transmitter, because of
the low attenuation of this type of line. (3) Coaxial line (usually RG -8 /U with a 52 -ohm
characteristic impedance) is widely used in
v -h -f work and also on the lower frequencies
where the feed line must run underground or
through the walls of a building. Coaxial line
also is of assistance in TVI reduction since
the r -f field is entirely enclosed within the
line. Molded 75 -ohm line is sometimes used
to feed a doublet antenna, but the doublet has
been largely superseded by the folded -dipole
antenna fed by 300 -ohm ribbon or tubular line
when an antenna for a single band is required.
discussed earlier,
standing waves on the antenna transmission line, in the transmitting case,
are a result of reflection from the point where
the feed line joins the antenna system. The
magnitude of the standing waves is deterStanding Waves
As was
mined by the degree of mismatch between the
characteristic impedance of the transmission
line and the input impedance of the antenna
system. When the feed -point impedance of the
antenna is resistive and of the same value as
the characteristic impedance of the feed line,
standing waves will not exist on the feeder.
It may be well to repeat at this time that there
is no adjustment which can be made at the
pedance of the transmission line is transformed gradually into a higher value by the
fanned -out Y portion of the feeders, and the
Y portion is tapped on the antenna at points
where the antenna impedance is a compromise
between the impedance at the ends of the Y
and the impedance of the unfanned portion of
the line.
The constants of the system are rather critical, and the antenna must resonate at the
operating frequency in order to minimize standing waves on the line. Some slight readjustment of the taps on the antenna is desirable,
if appreciable standing waves persist in appearing on the line.
The constants for a doublet are determined
by the following formulas:
L feet
-
467.4
F megacycles
175
Dfeet
F megacycles
Efeet
147.6
F megacycles
Where L is antenna length; D is the distance in
from each end at which the Y taps on; E is the
height of the Y section.
Since these constants are correct only for a
600 -ohm transmission line, the spacing S of
the line must be approximately 75 times the
diameter of the wire used in the transmission
line. For no. 14 B & S wire, the spacing will
be slightly less than 5 inches. This system
should never be used on either its even or odd
harmonics, as entirely different constants are
required when more than a single half wavelength appears on the radiating portion of the
system.
Multi -Wire Doublets
When a doublet antenna
or the driven element in
consists of more than one wire or
tubing conductor the radiation resistance of
the antenna or array is increased slightly as a
result of the increase in the effective diameter
of the element. Further, if we split just one
an array
www.americanradiohistory.com
438
THE
Antennas and Antenna Matching
RADIO
VOLTAGE CURVES
40 MET R5
20vMETERS
IO METERS
\%
8
AN
TOS MSC -TOP
87 FT.
VIA
33.5
ro
!CwLtILMIw
wo wwL*oRa
FT.
APE, Z' HIGH
CENTER
ANTENNA WIRE
FEEDER
Figure 27
SINGLE- WIRE -FED ANTENNA FOR ALL BAND OPERATION
An antenna of this type for 40 -, 20- and 10meter operation would have a radiator 67
feet long, with the feeder tapped 11 feet off
center. The feeder can be 33, 66 or 99 feet
long. The some type of antenna for 80 -, 40 -,
20- and 10 -meter operation would have a
radiator 134 feet long, with the feeder topped
22 feet off center. The feeder can be either
66 or 132 feet long. This system should be
used only with those coupling methods which
provide good harmonic suppression.
JOA« ALL
a2 Om.
cODUL
wiAEST
SASE
L.wE
L«[
Figure 26
MECHANICAL CONSTRUCTION
METER DISCONE
OF
20-
pending upon the wire size and the point of
attachment to the antenna. The earth losses
are comparatively low over ground of good conductivity. Since the single wire feeder radiates,
it is necessary to bring it away from the antenna at right angle s to the antenna wire
for at least one -half the length of the antenna.
The correct point for best impedance match
on the fundamental frequency is not suitable
for harmonic operation of the antenna. In addition, the correct length of the antenna for
fundamental operation is not correct for harmonic operation. Consequently, a compromise
must be made in antenna length and point of
feeder connection to enable the single -wirefed antenna to operate on more than one band.
Such a compromise introduces additional reactance into the single wire feeder, and might
cause loading difficulties with pi- network
transmitters. To minimize this trouble, the
single wire feeder should be made a multiple
of 33 feet long.
Two typical single -wire -fed antenna systems are shown in figure 27 with dimensions
for multi -band operation.
22 -8
Matching Non -Resonant
Lines to the Antenna
Present practice in regard to the use of
transmission lines for feeding antenna systems
on the amateur bands is about equally divided
www.americanradiohistory.com
HANDBOOK
Frequency
Low
i
F
3.3
S
437
.x.o...
...........11
..1....
M.^¡E
S`4.0
D
Discone
.,I............
30
II
O
i
I I
II
l.s
CC
I I
m
I I
H
I
z 25
I I
e
Io
14
le
22
26
30
34
3e
42
4e
50
54
5e
FREQUENCY (Mc)
II
II
I
Figure 25
CURVE FOR A 13.2 MC. DISCONE
ANTENNA. SWR IS BELOW 1.5 TO
FROM
II
SWR
1
13.0 MC. TO 58 MC.
R
32 OHM COAXIAL
FEED LINE
DIMENSIONS
20,15.11,10,4 METERS
D. 12
L.
S=10-
R. is,
1e
H.15' 7.
15, 1.10,6 METERS
1
=e
S. e.
D
L= 12'
R
=12,
H =10.5
11,10.6,2 METERS
=e
D=
L=Ye-
S
R=ee
H=e'3^
Figure 24
DIMENSIONS OF LOW- FREQUENCY DISCONE ANTENNA FOR LOW FREQUENCY
CUTOFF AT 13.2 MC., 20.1 MC., AND
26 MC.
The Discone is a vertically polarized radiator, producing an omnidirectional pattern
similar to a ground plane. Operation on several amateur bands with low SWR on the coaxial feed line is possible. Additional information on L -F Discone by W2RYI in July,
1950 CQ magazine.
of the radials may be reduced to 25 feet. As
with all multi -band antennas that employ no
lumped tuned circuits, this antenna offers no
attenuation to harmonics of the transmitter.
When operating on the lower frequency band,
it would be wise to check the transmitter for
second harmonic emission, since this antenna
will effectively radiate this harmonic.
The Low - Frequency
Discone
it
discone antenna is
widely used on the v -h -f
bands, but until recently
The
has not been put to any great use on the
lower frequency bands. Since the discone is a
broad -band device, it may be used on several
harmonically related amateur bands. Size is
the limiting factor in the use of a discone, and
the 20 meter band is about the lowest practical frequency for a discone of reasonable dimensions. A discone designed for 20 meter
operation may be used on 20, 15, 11, 10 and
6 meters with excellent results. It affords a
good match to a 50 ohm coaxial feed system
on all of these bands. A practical discone antenna is shown in figure 24, with a SWR curve
for its operation over the frequency range of
13 -55 Mc. shown in figure 25. The discone
antenna radiates a vertically polarized wave
and has a very low angle of radiation. For
v -h -f work the discone is constructed of sheet
metal, but for low frequency work it may be
made of copper wire and aluminum angle
stock. A suitable mechanical layout for a low
frequency di scone is shown in figure 26.
Smaller versions of this antenna may be constructed for 15, 11, 10 and 6 meters, or for 11,
10, 6 and 2 meters as shown in the chart of
figure 24.
For minimum wind resistance, the top
"hat"
of the discone is constructed from three -quarter inch aluminum angle stock, the rods being
bolted to an aluminum plate at the center of
the structure. The tips of the rods are all connected together by lengths of no. 12 enamelled
copper wire. The cone elements are made of
no. 12 copper wire and act as guy wires for
the discone structure. A very rigid arrangement may be made from this design; one that
will give no trouble in high winds. A 4" x 4"
post can be used to support the discone structure.
The discone antenna may be fed by a length
of 50 -ohm coaxial cable directly from the transmitter, with a very low SWR on all bands.
The old favorite single -wirefed antenna system is quite
satisfactory for an impromptu all band antenna system. It is widely used
for portable installations and "Field Day"
contests where a simple, multi -band antenna
is required. A single wire feeder has a characteristic impedance of some 500 ohms, deThe Single -WireFed Antenna
www.americanradiohistory.com
436
THE
Antennas and Antenna Matching
of transmission line of any characteristic impedance into a feeder system such as this and
the impedanc' at the far end of the line will
be exactly the same value of impedance which
the half -was e line sees at its termination.
Hence this has been done in the antenna system shown in figure 22; an electrical half
wave of line has been inserted between the
feed point of the antenna and the 300 -ohm
transmission line to the transmitter.
The characteristic impedance of this additional half -wave section of transmission line
has been made about 715 ohms (no. 20 wire
spaced 6 inches), but since it is an electrical
half wave long at 7 Mc. and operates into a
load of 300 ohms at the antenna the 300 -ohm
Twin -Lead at the bottom of the half-wave section still sees an impedance of 300 ohms. The
additional half -wave section of transmission
line introduces a negligible amount of loss
since the current flowing in the section of line
is the same which would flow in a 300 -ohm
line at each end of the half -wave section, and
at all other points it is less than the current
which would flow in a 300 -ohm line since the
effective impedance is greater than 300 ohms
in the center of the half -wave section. This
means that the loss is less than it would be in
an equivalent length of 300 -ohm TwinLead
since this type of manufactured transmission
line is made up of conductors which are equivalent to no. 20 wire.
So we see that the added section of 715 -ohm
line has substantially no effect on the operation of the antenna system on the 7 -Mc. band.
However, when the flat top of the antenna is
operated on the 3.5-Mc. band the feed-point
impedance of the flat top is approximately
3500 ohms. Since the section of 715 -ohm transmission line is an electrical quarter-wave in
length on the 3.5-Mc. band, this section of
line will have the effect of transforming the
approximately 3500 ohms feed-point impedance of the antenna down to an impedance of
about 150 ohms which will result in a 2:1
standing -wave ratio on the 300 -ohm Twin -Lead
transmission line from the transmitter to the
antenna system.
The antenna system of figure 22 operates
with very low standing waves over the entire
7 -Mc. band, and it will operate with moderate
standing waves from 3500 to 3800 kc. in the
3.5-Mc. band and with sufficiently low standing -wave ratio so that it is quite usable over
the entire 3.5 -Mc. band.
This antenna system, as well as all other
types of multi -band antenna systems, must be
used in conjunction with some type of harmonic- reducing antenna tuning network even
though the system does present a convenient
impedance value on both bands.
RADIO
L
300 OHM OPEN -WIRE
TV TYPE LINE
I6Á -e0 METERS
TO'
L.
V=52'
60 -40 METERS
L =33'
V =2s'
/*P-6 RADIALS
5
OHM COAX IA
INE
Figure 23
THE MULTEE TWO -BAND ANTENNA
This compact antenna can be used with excellent results on 160/80 and 80/40 meters.
The feedline should be held as vertical as
possible, since it radiates when the antenna
is operated on its fundamental frequency.
The "Multee"
Antenna
An
antenna that works well
on 160 and 80 meters, or 80
and 40 meters and is sufficiently compact to permit erection on the average city lot is the W68CX Multee antenna,
illustrated in figure 23. The antenna evolves
from a vertical two wire radiator, fed on one
leg only. On the low frequency band the top
portion does little radiating, so it is folded
down to form a radiator for the higher frequency band. On the lower frequency band, the antenna acts as a top loaded vertical radiator,
while on the higher frequency band, the flattop does the radiating rather than the vertical
portion. The vertical portion acts as a quarter wave linear transformer, matching the 6000
ohm antenna impedance to the 50 ohm impedance of the coaxial transmission line.
The earth below a vertical radiator must be
of good conductivity not only to provide a low
resistance ground connection, but also to provide a good reflecting surface for the waves
radiated downward towards the ground. For
best results, a radial system should be installed beneath the antenna. For 160 -80 meter operation, six radials 50 feet in length,
made of no. 16 copper wire should be buried
just below the surface of the ground. While an
ordinary water pipe ground system with no
radials may be used, a system of radials will
provide a worthwhile increase in signal
strength. For 80 -40 meter operation, the length
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HANDBOOK
Multi -band Antennas
435
144
33'
OR
NV
LONG-
400 01.163 OPEN -WIRE
TV TYPE LINE
ANTENNA TUNER
OR
MATCH NOX
SMAL
LINE
Figure 21
MULTI - BAND ANTENNA USING FAN
DIPOLE TO LIMIT IMPEDANCE EXCURSIONS ON HARMONIC FREQUENCIES
-
wire spaced 4 to 6 inches the antenna system is sometimes called a center -fed zepp.
With this type of feeder the impedance at the
transmitter end of the feeder varies from about
70 ohms to approximately 5000 ohms, the same
as is encountered in an end -fed zepp antenna.
This great impedance ratio requires provision
for either series or parallel tuning of the feeders at the transmitter, and involves quite high
r -f voltages at various points along the feed
14
line.
If the feed line between the transmitter and
the antenna is made to have a characteristic
impedance of approximately 300 ohms the excursions in end -of- feeder impedance are greatly reduced. In fact the impedance then varies
from approximately 75 ohms to 1200 ohms.
With this much lowered impedance variation
it is usually possible to use series tuning on
all bands, or merely to couple the antenna directly to the output tank circuit or the harmonic reduction circuit without any separate
feeder tuning provision.
There are several practicable types of transmission line which can give an impedance of
approximately 300 ohms. The first is, obviously, 300 -ohm Twin -Lead. Twin -Lead of the receiving type may be used as a resonant feed
line in this case, but its use is not recommended with power levels greater than perhaps
150 watts, and it should not be used when
lowest loss in the transmission line is desired.
For power levels up to 250 watts or so, the
transmitting type tubular 300 -ohm line may be
used, or the open -wire 300 -ohm TV line may
be employed. For power levels higher than
this, a 4- wire transmission line, or a line
built of one -quarter inch tubing should be
used.
FOLDED -TOP
Figure 22
DUAL -BAND ANTENNA
Even when a 300 -ohm transmission line is
used, the end-of- feeder impedance may reach
a high value, particularly on the second harmonic of the antenna. To limit the impedance
excursions,, a two -wire flat -top may be employed for the radiator, as shown in figure 21.
The use of such a radiator will limit the impedance excursions on the harmonic frequencies of the antenna and make the operation of
the antenna matching unit much less critical.
The use of a two -wire radiator is highly recommended for any center -fed multi -band antenna.
Folded Flot -Top
Dual -Band Antenna
As has been mentioned
earlier, there is an increasing tendency among amateur operators to utilize rotary or fixed arrays
for the 14-Mc. band and those higher in frequency. In order to afford complete coverage
of the amateur bands it is then desirable to
have an additional system which will operate
with equal effectiveness on the 3.5 -Mc. and
7 -Mc. bands, but this low- frequency antenna
system will not be required to operate on any
bands higher in frequency than the 7 -Mc. band.
The antenna system shown in figure 22 has
been developed to fill this need.
This system consists essentially of an
open -line folded dipole for the 7 -Mc. band with
a special feed system which allows the antenna to be fed with minimum standing waves
on the feed line on both the 7 -Mc. and 3.5 -Mc.
bands. The feed -point impedance of a folded
dipole on its fundamental frequency is approximately 300 ohms. Hence the 300 -ohm Twin Lead shown in figure 22 can be connected directly into the center of the system for operation only on the 7 -Mc. band and standing waves
on the feeder will be very small. However, it
is possible to insert an electrical half -wave
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4
34
RADIO
THE
Antennas and Antenna Matching
-lao -
r
L.90' FOR /0 -40 IMETES
1551014 LIN[ If
fOOR
VIED FOR 1-2 /MI IMPEDANCE AT
TNt TRANSMITTER END OF TOE
iP
BANDS
LI
La
TOPE OF
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LiNt if AP02.
L2
1200 ONMf
50
orHc
OPERATION
PARALLEL
11.
SS
PALACELI
250 LIMP
n5
Figure
A
19
TWO -BAND MARCONI ANTENNA
160 -80 METER OPERATION
105
FOR
7
!di
RS
T
Since this antenna type is an unbalanced radiating system, its use is not recommended with
high -power transmitters where interference to
broadcast listeners is likely to be encountered.
The r-f voltages encountered at the end of
zepp feeders and at points an electrical half
wave from the end are likely to be quite high.
Hence the feeders should be supported an adequate distance from surrounding objects and
sufficiently in the clear so that a chance encounter between a passerby and the feeder is
unlikely.
The coupling coil at the transmitter end of
the feeder system should be link coupled to
the output of the low -pass TVI filter in order
to reduce harmonic radiation.
The Two -Band
Marconi Antenna
A
three- eighths wavelength
antenna may be
operated on its harmonic
Marconi
frequency, providing good two band performance from a simple wire. Such an arrangement
for operation on 160 -80 meters, and 80 -40 meters is shown in figure 19. On the fundamental
(lowest) frequency, the antenna acts as a
three- eighths wavelength series -tuned Marconi.
On the second harmonic, the antenna is a current -fed three -quarter wavelength antenna operating against ground. For proper operation,
the antenna should be resonated on its second
harmonic by means of a grid-dip oscillator to
the operating frequency most used on this particular band. The Q of the antenna is relatively
low, and the antenna will perform well over a
frequency range of several hundred kilocycles.
The overall length of the antenna may be
varied slightly to place its self- resonant frequency in the desired region. Bends or turns
in the antenna tend to make it resonate higher
in frequency, and it may be necessary to
lengthen it a bit to resonate it at the chosen
frequency. For fundamental operation, the
series condenser is inserted in the circuit, and
the antenna may be resonated to any point in
the lower frequency band. As with any Marconi
n
MRCS
PA
PARALLEL
5ERIEs
Mc
MC
stain
PARALLEL
SI
S]
PA
14
MC
211
MC
PARALLEL
SI
1100 Ow.
1100 ON25
M
0.045
w
Tf 00uf
taw 0005
0021
A
75 OHMS
/200 OHMS
55
1200
1200
o02í
1200 ONMs
Mc
Mc
1200 ONMf
1100 OHMS
100
1200 OHMS
Haw
Ms
Ms
:00
CENTER-FED ANTENNA
DIMENSIONS
Figure 20
CENTER -FED MULTI
BAND ANTENNA
FOR
-
type antenna, the use of a good ground is essential. This antenna works well with transmitters employing coaxial antenna feed, since
its transmitting impedance on both bands is in
the neighborhood of 40 to 60 ohms. It may be
attached directly to the output terminal of such
transmitters as the Collins 32V and the Viking
H. The use of a low -pass TVI filter is of
course recommended.
For multi -band operation,
the center fed antenna is
without doubt the best
compromise. It is a balanced system on all
bands, it requires no ground return, and when
properly tuned has good rejection properties
for the higher harmonics generated in the transmitter. It is well suited for use with the various
multi -band 150 -watt transmitters that are currently so popular. For proper operation with
The Center -Fed
Multi -Band Antenna
these transmitters, an antenna tuning unit
must be used with the center-fed antenna. In
fact, some sort of tuning unit is necessary for
any type of efficient, multi -band antenna. The
use of such questionable antennas as the "offcenter fed15 doublet is an invitation to TVI
troubles and improper operation of the transmitter. A properly balanced antenna is the
best solution to multi -band operation. When
used in conjunction with an antenna tuning
unit, it will perform with top efficiency on all
of the major amateur bands.
Several types of center -fed antenna systems
are shown in figure 20. If the feed line is made
up in the conventional manner of no. 12 or no.
HANDBOOK
`[ rant+ NH...
Multi -band Antennas
a
6'
L
I11
L N'
L 411'
100 OMM
O
433
FEEDER SPREADERS
'OR 3310 NC AND 7114 NC
rpm 7I0O NC AND 1310 NC.
rpA I4100
NC. AND
a=
MC
N*
'"ITEM
600 A LINE
THE
Figure 15
THREE -QUARTER WAVE
DOUBLET
This
antenna
arrangement
FOLDED
SHORTED END
600 OHM LINE
TO TRANSMITTER
L.
will give very
the fundamental frequency
and
with
1
.
- 496
S FT
effective radiator on the second harmonic but
the pattern of radiation will be different from
that on the fundamental, and the standing -wave
ratio on the feed line will be greater. The flat
top of the antenna must be made of open wire
rather than ribbon or tubular line.
For greater operating convenience, the shorting switch may be replaced with a section of
transmission line. If this transmission line is
made one -quarter wavelength long for the fundamental frequency, and the free end of the
line is shorted, it will act as an open circuit
across the center insulator. At the second harmonic, the transmission line is one -half wavelength long, and reflects the low impedance
of the shorted end across the center insulator.
Thus the switching action is automatic as the
frequency of operation is changed. Such an
installation is shown in figure 16.
The end -fed Hertz antenna
shown in figure 17 is not as
effective a radiating system as
Figure 16
AUTOMATIC BANDSWITCHING STUB FOR
THE THREE -QUARTER WAVE FOLDED
DOUBLET
The antenna of Figure 15 may be used with
a shorted stub line in place of the switch
normally used for second harmonic operation.
types, but it is particularit is desired to install an
for a test, or for field -day
of the radiator should be
clear as possible. In any
event at least three quarters of the total wire
length should be in the clear. Dimensions for
optimum operation on various amateur bands
are given in addition in figure 17.
many other antenna
ly convenient when
antenna in a hurry
work. The flat top
as high and in the
The end- f ed Zepp has long
been a favorite for multi -band
operation. It is shown in figure 18 along with recommended dimensions
for operation on various amateur band groups.
The End -Fed
Zepp
-
LI
r01÷ .1ta
3.5 AND
7
MC.
3.9 MC. AND 26 MC.
Ll36
a.1 MC
SS
MC
L= 137'
3,5
11C
L136
1
LO
20'
7
OR1
Sr1IAD[RS
all
MC
MC
SILO.-
L1
TTn Dr
1n
a
$10,01
MRALL[L
137
n
$1
14
BANDS
3.3, 7, 14 AND 26 MC.
3.5, 7 AND 14 MC.
MC
FROM
xMTR.
17
END-FED ZEPP
RECOMMENDED LENGTHS FOR THE END FED HERTZ
www.americanradiohistory.com
FIGURE 18
TIMING
110610/
M1ALL11L
/ARALLIL
M1
MIES
LINK
Figure
FT
the
switch closed on twice frequency.
The End -Fed
Hertz
67 FT WHEN ANTENNA IS 195 FT.
- 96 Fr.
L' 33 FT
L ' 6
satisfactory operation with a 600 -ohm feed
line for operation with the switch open on
-
-- -
-
432
THE
Antennas and Antenna Matching
3
RADIO
-
ANTENNA
eO METER7
L13sC
4001lUr
PNENOUC !LOCKS
SEE
3.5
3.1
3.7
30
3
rIC.12
40
SWR
C200YLr
Figure 13
CURVE OF 80 -METER BROAD-BAND
INNUR CONDUCTOR NOT USED
SEE FIG.12 FOR CONNECTION
DIPOLE
52 OHM COAXIAL LINE
Figure
ohms. The ground losses are now reduced by
a factor of 4. In addition, the antenna may be
directly fed from a 50 -ohm coaxial line, or directly from the unbalanced output of a pi- network transmitter.
Since a certain amount of power may still
be lost in the ground connection, it is still of
greatest importance that a good, low resistance ground be used with this antenna.
Shown in figures 11 and 12
are broad -band dipoles for
the 40 and 80 meter amateur
bands, designed by Collins
Radio Co. for use with the Collins 32V -3 and
KW -1 transmitters. These fan -type dipoles
have excellent broad-band response, and are
designed to be fed with a 52 -ohm unbalanced
coaxial line, making them suitable for use with
many of the other modem transmitters, such
as the Barker and Williamson 5100, Johnson
Ranger, and Viking. The antenna system consists of a fan -type dipole, a balun matching
section, and a suitable coaxial feedline. The
Q of the half -wave 80 meter doublet is lowered by decreasing the effective length -todiameter ratio. The frequency range of operation of the doublet is increased considerably
by this change. A typical SWR curve for the
80 meter doublet is shown in figure 13.
The balanced doublet is matched to the unbalanced coaxial line by the one -quarter wave
balun. If desired, a shortened balun may be
used (figure 14). The short balun is capacity
loaded at the junction between the balun and
the broad-band dipole.
The Collins
Brood -bond
Dipole System
22 -7
Multi -Band Antennas
The availability of a multi -band antenna is
great operating convenience to an amateur
station. In most cases it will be found best to
install an antenna which is optimum for the
band which is used for the majority of the
a
M
L7'3-
4.0
FREQUENCY (MC)
14
SHORT BALUN FOR 40 AND 80 METERS
available operating time, and then to have an
additional multi -band antenna which may be
pressed into service for operation on another
band when propagation conditions on the most
frequently used band are not suitable. Most
amateurs use, or plan to install, at least one
directive array for one of the higher- frequency
bands, but find that an additional antenna
which may be used on the 3.5-Mc. and 7.0 -Mc.
band, or even up through the 28 -Mc. band is
almost indispensable.
The choice of a multi -band antenna depends
upon a number of factors such as the amount
of space available, the band which is to be
used for the majority of operating with the antenna, the radiation efficiency which is desired, and the type of antenna tuning network
to be used at the transmitter. A number of
recommended types are shown in the next
pages.
The
3í -Wave
Folded Doublet
Figure
15
type which
be
very
shows an antenna
will be found to
effective when a
moderate amount of space is available, when
most of the operating will be done on one band
with occasional operation on the second harmonic. The system is quite satisfactory for
use with high -power transmitters since a 600 ohm non -resonant line is used from the antenna to the transmitter and since the antenna
system is balanced with respect to ground.
With operation on the fundamental frequency
of the antenna where the flat top is % wave
long the switch SW is left open. The system
affords a very close match between the 600ohm line and the feed point of the antenna.
Kraus has reported a standing -wave ratio of
approximately 1.2 to 1 over the 14 -Mc. band
when the antenna was located approximately
one -half wave above ground.
For operation on the second harmonic the
switch SW is closed. The antenna is still an
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HANDBOOK
1
110'
44.9
r
1ldllllllr
FOR
11.10.10c
FOR
DETAIL SEE FIG. A
PHENOLIC BLOCK 2XI.5XC
WRAP CABLES AND BLOCK
WITH SCOTCH ELECTRICALTAPE
SPACE BLOCKS 0' APART
ALONG BALUN
+j//
2 X 1.3 X 0.5
ENOLIC BLOCK
/ //
//%////
////
///
431
411111110
THE TWO W IRES MAY BE
SPREAD EITHER HORIZONTALLYOR VERTICALLY.
WRAP CABLES AND BLOCK
ITN SCOTCH ELECTRICAL T
SPACE BLOCKS B'APART
ALONG BALUN
i
iW.N.I
'-11.11.IuIrI
114
OS-
rI.Y
0.5W
LS-
FIGURE A
FIGURE B
FIGURE A
CUT OFF SHIELD AND OUTER
JACKET AS SHOWN. ALLOW
DIELECTRIC TO E %TEND PART
WAY TO OTHER CABLE. COVER
ALL EXPOSED SHIELD AND
DIELECTRIC ON BOTH CABLES
WITH A CONTINUOUS WRAPPING OF SCOTCH ELECTRICAL
TAPE TO EXCLUDE MOISTURE.
B
KEEP BALUN AT LEAST
CLEAR
OF GROUND AND OTHER OBJECTS.
FOR
DETAIL SEE FIG A
-
Antennas
Conserving
Space
DETAIL SEE FIGURE
B
52 OHM RG-8/U, ANY LENGTH
REMOVE OUTER JACKET
FROM A SHORT LENGTH OF
CABLE AS SHOWN HERE.
UNBRAID THE SHIELD OF
COAX
CUT OFF THE
DI-
ELECTRIC AND INNER CONDUCTOR FLUSH WITH THE
OUTER JACKET. DO HOT CUT
THE SHIELD. WRA SHIELD
DIELECTRIC ON
CONNECTION. BEING VERY
CAREFUL NOT TO DAMAGE
THE DIELECTRIC MATERIAL.
HOLD CABLE O STRAIGHT
WHILE SOLDERING. COVER
THE AREA WITH A CONTINUOUS WRAPPING OF SCOTCH
ELECTRICAL TAPE. NO CONNECTION TO INNER CONDUC-
KEEP BALUN AT LEAST
OF COAX C AROUND SHIELD
OF COAX D. SOLDER THE
FIGURE B
REMOVE OUTER JACKET
FROM A SHORT LENGTH OF
CABLE AS SHOWN HERE.
UNBRAID THE SHIELD OF
COAX C CUTOFF THE DIELECTI(IC AND INNER CON DUCTOR FLUSH WITH THE
OUTER JACKET. DO NOT CUT
THE SHIELD. WRAP SHIELD
OF COAX C AROUND SHIELD
OF COAX D. SOLDER THE
CONNECTION. BEING VERY
CAREFUL NOT TO DAMAGE
THE DIELECTRIC MATERIAL.
HOLD CABLE D STRAIGHT
WHILE SOLDERING. COVER
CUT OFF SHIELD AND OUTER
JACKET AS SHOWN. ALLOW
DIELECTRIC TO EXTEND PART
WAY TO OTHER CABLE. COVER
ALL EXPOSED SHIELD AND
BOTH
CABLES
WRAP-
PING OF SCOTCH ELECTRICAL
TAPE TO EXCLUDE MOISTURE.
OF GROUND AND OTHER
FOR
B-
LEAR
OBJECTS.
DETAIL SEE FIGURE
B
THE AREA WITHACONTIN-
52 OHM
UOUS WRAPPING OF SCOTCH
ELECTRICAL TAPE. N0 CONNEC T ION TO INNER CONDUCTORS.
RD -B /U, ANY LENGTH
TORS.
DIMENSIONS SHOWN NERE ARE FOR THE 40 METER BAND. THIS ANTENNA MAY BE BUILT FOR OTHER BANDS BY US/Ni DIMENSIONS THAT
ARE MULTIPLES OR SUBMUL TIPLES OF THE DIMENSIONS SHOWN.
BALUN SPACING IS S. ON ALL BANDS.
Figure 11
HALF -WAVE ANTENNA WITH QUARTER WAVE
UNBALANCED TO BALANCED
TRANSFORMER (BALUN) FEED SYSTEM
FOR 40-METER OPERATION
sions in terms of frequency are given on the
drawing. An antenna of this type is 93 feet
long for operation on 3600 kc. and 86 feet long
for operation on 3900 kc. This type of antenna
has the additional advantage that it may be
operated on the 7 -Mc. and 14 -Mc. bands, when
the flat top has been cut for the 3.5 -Mc. band,
simply by changing the position of the shorting bar and the feeder line on the stub.
A sacrifice which must be made when using
a shortened radiating system, as for example
the types shown in figure 9, is in the bandwidth of the radiating system. The frequency
range which may be covered by a shortened
antenna system is approximately in proportion
to the amount of shortening which has been
employed. For example, the antenna system
shown in figure 9C may be operated over the
range from 3800 kc. to 4000 kc. without serious standing waves on the feed line. If the
b
DIMENSIONS SHOWN HERE ARE FOR THE
METER BAND. THIS ANTENNA MAY BE BUILT FOR OTHER BANDS BY USINE DIMENSIONS THAT
ARE MULTIPLES OR SUBMUL TIPLES OF THE DIMENSIONS SHOWN.
BALUN SPACING IS /.5. ON ALL BANDS.
Figure 12
BROADBAND ANTENNA WITH QUARTER WAVE
UNBALANCED
TO BALANCED
TRANSFORMER (BALUN) FEED SYSTEM
FOR 80 -METER OPERATION
antenna had been made full length it would
be possible to cover about half again as much
frequency range for the same amount of mismatch on the extremes of the frequency range.
The Twin -Lead
Marconi Antenna
Much of the power
loss in
the Marconi antenna is a result of low radiation resistance and high ground resistance. In some
cases, the ground resistance may even be
be higher than the radiation resistance, causing a loss of 50 per cent or more of the transmitter power output. If the radiation resistance
of the Marconi antenna is raised, the amount
of power lost in the ground resistance is proportionately less. If a Marconi antenna is made
out of 300 ohm TV -type ribbon line, as shown
in figure 10, the radiation resistance of the
antenna is raised from a low value of 10 or 15
ohms to a more reasonable value of 40 to 60
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430
THE
Antennas and Antenna Matching
.. .,.
AT
Lo..,T
RADIO
/MM.,
300 OHM -RIBBON- LINE
YrLCGaa
areeoaXa
WIRES SHORTED TOGETHER AT END
52A COAXIAL
TEED LINE
Figure
10
TWIN -LEAD MARCONI ANTENNA FOR THE
80 AND 160 METER BANDS
Figure 9
THREE EFFECTIVE SPACE CONSERVING
ANTENNAS
The arrangements shown at (A) and (B) are
satisfactory where resonant feed line can be
used. However, non- resonant 75 -ohm feed
line may be used in the arrangement at (A)
when the dimensions in wavelengths are as
shown. In the arrangement shown at (B) low
standing waves will be obtained on the feed
line when the overall length of the antenna
is a half wave. The arrangement shown at
(C) may be tuned for any reasonable length
of flat top to give a minimum of standing
waves on the transmission line.
quarter wavelength can be lengthened electrically by means of a series loading coil, and
used as a quarter -wave Marconi. However, if
the wire is made shorter than approximately
one -eighth wavelength, the radiation resistance will be quite low. This is a special problem in mobile work below about 20 -Mc.
22 -6
Space -Conserving
Antennas
In many cases it is desired to undertake a
considerable amount of operation on the 80meter or 40 -meter band, but sufficient space
is simply not available for the installation of
a half-wave radiator for the desired frequency
of operation. This is a common experience of
apartment dwellers. The shortened Marconi
antenna operated against a good ground can
be used under certain conditions, but the shortened Marconi is notorious for the production
of broadcast interference, and a good ground
connection is usually completely unobtainable
in an apartment house.
Essentially, the problem in producing an
antenna for lower frequency operation in restricted space is to erect a short radiator
which is balanced with respect to ground and
which is therefore independent of ground for
its operation. Several antenna types meeting
this set of conditions are shown in figure 9.
Figure 9A shows a conventional center -fed
doublet with bent -down ends. This type of antenna can be fed with 75-ohm Twin -Lead in the
center, or it may be fed with a resonant line
for operation on several bands. The overall
length of the radiating wire will be a few per
cent greater than the normal length for such
an antenna since the wire is bent at a position intermediate between a current loop and
a voltage loop. The actual length will have to
be determined by the cut-and -try process because of the increased effect of interfering objects on the effective electrical length of an
antenna of this type.
Figure 9B shows a method for using a two wire doublet on one half of its normal operating frequency. It is recommended that spaced
open conductor be used both for the radiating
portion of the folded dipole and for the feed
line. The reason for this recommendation lies
in the fact that the two wires of the flat top
are not at the same potential throughout their
length when the antenna is operated on one half frequency. Twin -Lead may be used for
the feed line if operation on the frequency
where the flat top is one -half wave in length
is most common, and operation on one -half frequency is infrequent. However, if the antenna
is to be used primarily on one -half frequency
as shown, it should be fed by means of an
open-wire line. If it is desired to feed the antenna with a non -resonant line, a quarter -wave
stub may be connected to the antenna at the
points X, X in figure 9B. The stub should be
tuned and the transmission line connected to
it in the normal manner.
The antenna system shown in figure 9C may
be used when not quite enough length is available for a full half-wave radiator. The dimen-
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HANDBOOK
Marconi
Antenn a
LOADING
COILS
Figure
429
`MAT
a
7
8
LOADING THE
MARCONI ANTENNA
The various loading systems
are discussed in the accom-
panying text.
O
current flows through a r e s i s t o r, or if the
ground itself presents some resistance, there
will be a power loss in the form of heat. Improving the ground connection, therefore, provides a definite means of reducing this power
loss, and thus increasing the radiated power.
The best possible ground consists of as
many wires as possible, each at least a quarter wave long, buried just below the surface
of the earth, and extending out from a common
point in the form of radials. Copper wire of
any size larger than no. 16 is satisfactory,
though the larger sizes will take longer to disintegrate. In fact, the radials need not even
be buried; they may be supported just above
the earth, and insulated from it. This arrangement is called a counterpoise, and operates
by virtue of its high capacitance to ground.
If the antenna is physically shorter than a
quarter wavelength, the antenna current is
higher, due to lower radiation resistance. Consequently, the power lost in resistive soil is
greater. The importance of a good ground with
short, inductive -loaded Marconi radiators is,
therefore, quite obvious. With a good ground
system, even very short (one- eighth wavelength) antennas can be expected to give a
high percentage of the efficiency of a quarterwave antenna used with the same ground system. This is especially true when the short
radiator is top loaded with a high Q (low loss)
coil.
Water-Pipe
Grounds
Water pipe, because of its corn -
paratively large surface and cross
section, has a relatively low r -f
resistance. If it is possible to attach to a
junction of several water pipes (where they
branch in several directions and run for some
distance under ground), a satisfactory ground
connection will be obtained. If one of the
pipes attaches to a lawn or garden sprinkler
system in the immediate vicinity of the antenna, the effectiveness of the system will approach that of buried copper radials.
The main objection to water-pipe grounds
©
©
0
0
0
is the possibility of high resistance joints in
the pipe, due to the "dope" put on the coupling threads. By attaching the ground wire
to a junction with three or more legs, the possibility of requiring the main portion of the
r -f current to flow through a high resistance
connection is greatly reduced.
The presence of water in the pipe adds
nothing to the conductivity; therefore it does
not relieve the problem of high resistance
joints. Bonding the joints is the best insurance, but this is, of course, impracticable
where the pipe is buried. Bonding together
with copper wire the various water faucets
above the surface of the ground will improve
the effectiveness of a water -pipe ground system hampered by high -resistance pipe couplings.
antenna is an odd
of electrical quarter
waves long (usually only one
quarter wave in length), and is always resonated to the operating frequency. The correct
loading of the final amplifier is accomplished
by varying the coupling, rather than by detuning the antenna from resonance.
Physically, a quarter -wave Marconi may be
made anywhere from one - eighth to three-eighths
wavelength overall, meaning the total length of
the antenna wire and ground lead from the end
of the antenna to the point where the ground
lead attaches to the junction of the radials or
counterpoise wires, or where the water pipe
enters the ground. The longer the antenna
is made physically, the lower will be the current flowing in the ground connection, and the
greater will be the overall radiation efficiency.
However, when the antenna length exceeds
three -eighths wavelength, the antenna becomes difficult to resonate by means of a
series capacitor, and it begins to take shape
as an end -fed Hertz, requiring a method of
feed such as a pi network.
A radiator physically much shorter than a
Marconi
Marconi
A
Dimensions
number
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428
THE
Antennas and Antenna Matching
RADIO
used for the radiator. Such an ant e nn a is
shown in figure 6. The loaded ground-plane
tends to have a rather high operating Q and
operates only over a narrow band of frequencies. An operating range of about 100 kilo-
cycles with
a low SWR is possible on 80 meters. Operation over a larger frequency range
is possible if a.higher standing wave ratio is
tolerated on the transmission line. The radiation resistance of a loaded 80 -meter groundplane is about 15 ohms. A quarter wavelength
(45 feet) of 52 -ohm coaxial line will act as an
efficient feed line, presenting a load of approximately 180 ohms to the transmitter.
22 -5
COAX. PROM TRANS.
Figure 7
FEEDING A QUARTER -WAVE MARCONI
ANTENNA
When an open -wire line is to be used, it may
be link coupled to o series- resonant circuit
between the bottom end of the Marconi and
ground, as of (A). Alternatively, a reasonably good impedance match may be obtained
between 52 -ohm coaxial line and the bottom
of a resonant quarter -wave antenna, as illustrated at (B) above.
The Marconi
Antenna
A grounded quarter-wave Marconi antenna,
widely used on frequencies below 3 Mc., is
sometimes used on the 3.5-Mc. band, and is
also used in v -h -f mobile services where a
compact antenna is required. The Marconi type
antenna allows the use of half the length of
wire that would be required for a half -wave
Hertz radiator. The ground acts as a mirror,
in effect, and takes the place of the additional
quarter -wave of wire that would be required
to reach resonance if the end of the wire were
not returned to ground.
The fundamental practical form of the Marconi antenna system is shown in figure 7.
Other Marconi antennas differ from this type
primarily in regard to the method of feeding
the energy to the radiator. The feed method
shown in figure 7B can often be used to advantage, particularly in mobile work.
Variations on the basic Marconi antenna
are shown in the illustrations of figure 8. Figures 8B and 8C show the "L" -type and "T "type Marconi antennas. These arrangements
have been more or less superseded by the toploaded forms of the Marconi antenna shown in
figures 8D, 8E, and 8F. In each of these latter three figures an antenna somewhat less
than one quarter wave in length has been
loaded to increase its effective length by the
insertion of a loading coil at or near the top
of the radiator. The arrangement shown at figure 8D gives the least loading but is the most
practical mechanically. The system shown at
figure 8E gives an intermediate amount of
loading, while that shown at figure 8F, utilizing a "hat" just above the loading coil, gives
the greatest amount of loading. The object of
all the top -loading methods shown is to produce an increase in the effective length of
the radiator, and thus to raise the point of
maximum current in the radiator as far as pos-
sible above ground. Raising the maximum -current point in the radiator above ground has
two desirable results: The percentage of low angle radiation is increased and the amount of
ground current at the base of the radiator is
reduced, thus reducing the ground losses.
To estimate whether a loading coil will
probably be required, it is necessary only to
note if the length of the antenna wire and
ground lead is over a quarter wavelength; if
so, no loading coil is needed, provided the
series tuning capacitor has a high maximum
capacitance.
Amateurs primarily interested in the higher
frequency bands, but who like to work 80 meters occasionally, can usually manage to resonate one of their antennas as a Marconi by
working the whole system, feeders and all,
against a water pipe ground, and resorting to
a loading coil if necessary. A high- frequencyrotary, zepp, doublet, or single- wire -fed antenna will make quite a good 80 -meter Marconi
if
high and in the clear, with a rather Long
feed line to act as a radiator on 80 meters.
Where two-wire feeders are used, the feeders
should be tied together for Marconi operation.
Importance of
Ground Connection
With a quarter -wave antenna and a ground, the antenna current generally is
measured with a meter placed in the antenna
circuit close to the ground connection. If this
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HANDBOOK
Vertical Antennas
427
LOADING COIL
APROXI MAYFLY D! TURNS
RIZ WIRE, .S" DIAMETER
AND
RADIALS EACH
la
ground is an effective transmitting antenna for
low-angle radiation, where ground conditions
in the vicinity of the antenna are good. Such
an antenna is not good for short -range sky wave communication, such as is the normal
usage of the 3.5 -Mc. amateur band, but is excellent for short-range ground-wave communication such as on the standard broadcast band
and on the amateur 1.8 -Mc. band. The vertical
antenna normally will cause greater BO than
an equivalent horizontal antenna, due to the
much greater ground -wave field intensity. Also, the vertical antenna is poor for receiving
under conditions where man -made interference
is severe, since such interference is predominantly vertically polarized.
Three ways of feeding a half -wave vertical
antenna from an untuned transmission line are
illustrated in figure 4. The J -fed system shown
in figure 4A is obviously not practicable except on the higher frequencies where the extra length for the stub may easily be obtained.
However, in the normal case the ground-plane
vertical antenna is to be recommended over
the J -fed system for high frequency work.
22 -4
An
The Ground Plane Antenna
effective low angle radiator for any ama-
EACH
52 OHM COAXIAL LINE
45 FEET LONG
VERTICAL WHIP
Figure 5
THE LOW -FREQUENCY GROUND PLANE
ANTENNA
The radials o f the ground plane antenna
should lie in a horizontal plane, although
slight departures from this caused by nearby
objects is allowable. The whip may be
mounted on a short post, or on the roof of a
building. The wire radials may slope downwards towards their tips, acting as guy
wires for the installation.
FOOT LONG
RADIALS
52 OHM COAXIAL LINE,
CENTER CONDUCTOR CONNECTS
TO
I
Figure
6
80 METER LOADED GROUND
PLANE
ANTENNA
Number of turns in loading coil to be adjusted
until antenna system resonates at desired
frequency in 80 meter band.
teur band is the ground-plane antenna, shown
in figure 5. So called because of the radial
ground wires, the ground -plane antenna is not
affected by soil conditions in its vicinity due
to the creation of an artificial ground system
by the radial wires. The base impedance of
the ground plane is of the order of 30 to 35
ohms, and it may be fed with 52 -ohm coaxial
line with only a slight impedance mis -match.
For a more exact match, the ground-plane antenna may be fed with a 72 -ohm coaxial line
and a quarter-wave matching section made of
52 -ohm
coaxial line.
The angle of radiation of the ground -plane
antenna is quite low, and the antenna will be
found less effective for contacts under 1000
miles or so on the 80 and 40 meter bands than
a high angle radiator, such as a dipole. However, for DX contacts of 1000 miles or more,
the ground -plane antenna will prove to be
highly effective.
The 80 -Meter
Loaded
Ground -Plane
A vertical antenna of 66 feet
in height presents quite a problern on a small lot, as the supporting guy wires will tend to
take up quite a large portion of the lot. Under
such conditions, it is possible to shorten the
length of the vertical radiator of the ground plane by the inclusion of a loading coil in the
vertical whip section. The ground-plane antenna may be artificially loaded in this manner so that a 25 -foot vertical whip may be
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426
THE
Antennas and Antenna Matching
RADIO
462
FMc
tDZ
FNC.
I
--
'
o.A.:
oaPaii
1
FED
VERTICAL
300 -OHM RIBBON
STVB-FED
VERTICAL
©
L-C
FED
VERTICAL
404
FNc
300 -ONM RIBBON
30
FMC.
FOLDED
Figure 3
DIPOLE WITH SHORTING
STRAPS
The
impedance
match and bandwidth char-
acteristics ofa folded dipole maybe improved
by shorting the two wires of the ribbon a distance out from the center equal to the velocity factor of the ribbon times the half -length
of the dipole as shown at (A). An alternative
arrangement with bent down ends for space
conservation is illustrated at
(13).
times over the radiation resistance of the element, have both contributed to the frequent
use of the multi -wire radiator as the driven
element in a parasitic antenna array.
Delta-Matched
Doublet and
Standard Doublet
These two types of radiating elements are shown in
figure 2L and figure 2M. The
delta- matched doublet is
described in detail in Section 22 -8 of this
chapter. The standard doublet, shown in figure 2M, is fed in the center by means of 75ohm Twin -Lead, either the transmitting or the
receiving type, or it may be fed by means of
twisted -pair feeder or by means of parallel wire lamp -cord. Any of these types of feed
line will give an approximate match to the
center impedance of the dipole, but the 75ohm Twin -Lead is far to be preferred over the
other types of low -impedance feeder due to
the much lower losses of the polyethylene -
dielectric transmission line.
The coaxial- cable -fed doublet shown in figure 2N is a variation on the system shown in
figure 2M. Either 52 -ohm coaxial cable or 75ohm coaxial cable may be used to feed the
center of the dipole, although the 75 -ohm type
Figure 4
HALF -WAVE VERTICAL ANTENNA SHOWING ALTERNATIVE METHODS OF FEED
will give a somewhat better impedance match
at normal antenna heights. Due to the asymmetry of the coaxial feed system difficulty
may be encountered with waves traveling on
the outside of the coaxial cable. For this reason the use of Twin -Lead is normally to be
preferred over the use of coaxial cab e for
feeding the center of a half-wave dipole.
1
Off- Center
Fed Doublet
shown in figure
2(0) is sometimes used to
The system
feed a half -wave dipole, especially when it is desired to use the same antenna on a number of harmonically-related frequencies. The feeder wire (no. 14 enamelled
wire should be used) is tapped a distance of
14 per cent of the total length of the antenna
either side of center. The feeder wire, operating against ground for the return current, has
an impedance of approximately 600 ohms. The
system works well over highly conducting
ground, but will introduce rather high losses
when the antenna is located above rocky or
poorly conducting soil. The off -center fed antenna has a further disadvantage that it is
highly responsive to harmonics fed to it from
the transmitter.
The effectiveness of the antenna system in
radiating harmonics is of course an advantage
when operation of the antenna on a number of
frequency bands is desired. But it is necessary to use a harmonic filter to insure that
only the desired frequency is fed from the
transmitter to the antenna.
22 -3
The Half -Wave
Vertical Antenna
The half-wave vertical antenna with its bottom end from 0.1 to 0.2 wavelength a
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bo v e
THE
Multi -wire Doublets
RADIO
series with the antenna coil or in parallel
with it. A series tuning c a p a c i tor can be
placed in series with one feeder leg without
unbalancing the system.
The tuned -doublet antenna is shown in figure 2D. The antenna is a current-fed system
when the radiating wire is a half wave long
electrically, or when the system is operated
on its odd harmonics, but becomes a voltage fed radiator when operated on its even harin
monics.
The antenna has a different radiation pattern when operated on its harmonics, as would
be expected. The arrangement used on the
second harmonic is better known as the Franklin colinear array and is described in Chapter
Twenty- three. The pattern is similar toa 1,j-wave
dipole except that it is sharper in the broadside direction. On higher harmonics of operation there will be multiple lobes of radiation
from the system.
Figures 2E and 2F show alternative arrangements for using an untuned transmission line
between the transmitter and the tuned-doublet
radiator. In figure 2E a half -wave shorted line
is used to resonate the radiating system,
while in figure 2F a quarter-wave open line is
utilized. The adjustment of quarter -wave and
half -wave stubs is discussed in Section 19 -8.
Doublets with
Quarter -Wave
Transformers
The average value of feed impedance for a center -fed halfwave doublet is 75 ohms. The
actual value varies with height
and is shown in Chapter Twenty-one. Other
methods of matching this rather low value of
impedance to a medium -impedance transmission line are shown in (G), (H), and (I) of figure 2. Each of these three systems uses a
quarter -wave transformer to accomplish the
impedance transformation. The only difference
between the three systems lies in the type of
transmission line used in the quarter-wave
transformer. (G) shows the Johnson Q system
whereby a line made up of 1/2-inch dural tubing
is used for the low- impedance linear transformer. A line made up in this manner is frequently called a set of Q bars. Illustration
(H) shows the use of a four-wire line as the
linear transformer, and (I) shows the use of a
piece of 150 -ohm Twin -Lead electrically 1/2wave in length as the transformer between the
center of the dipole and a piece of 300 -ohm
Twin -Lead. In any case the impedance of the
quarter -wave transformer will be of the order
of 150 to 200 ohms. The use of sections of
transmission line as linear transformers is
discussed in detail in Section 22 -8.
Multi -Wire
Doublets
alternative method for increasing the feed-point impedance of a
dipole so that a medium -impedAn
425
ance transmission line may be used is shown
in figures 2J and 2K. This system utilizes
more than one wire in parallel for the radiating
element, but only one of the wires is broken
for attachment of the feeder. The most common arrangement uses two wires in the flat
top of the antenna so that an impedance multiplication of four is obtained.
The antenna shown in figure 2J is the socalled Twin -Lead folded dipole which is a
commonly used antenna system on the mediumfrequency amateur bands. In this arrangement
both the antenna and the transmission line to
the transmitter are constructed of 300 -ohm
Twin-Lead. The flat top of the antenna is
made slightly less than the conventional
length (462 /FMc, instead of 468 /FMc, for a
single -wire flat top) and the two ends of the
Twin -Lead are joined together at each end.
The center of one of the conductors of the
Twin -Lead flat top is broken and the two ends
of the Twin -Lead feeder are spliced into the
flat top leads. As a protection against moisture pieces of flat polyethylene taken from
another piece of 300 -ohm Twin -Lead may be
molded over the joint between conductors with
the aid of an electric iron or soldering iron.
Better bandwidth characteristics can be obtained with a folded dipole made of ribbon line
if the two conductors of the ribbon line are
shorted a distance of 0.82 (the velocity factor
of ribbon line) of a free -space quarter wavelength from the center or feed point. This procedure is illustrated in figure 3A. An alternative arrangement for a Twin -Lead folded
dipole is illustrated in figure 3B. This type of
half -wave antenna system is convenient for
use on the 3.5-Mc. band when the 116 to 132
foot distance required for a full half -wave is
not quite available in a straight line, since the
single -wire end pieces may be bent away or
downward from the direction of the main section of the antenna.
Figure 2K shows the basic type of 2 -wire
doublet or folded dipole wherein the radiating
section of the system is made up of standard
antenna wire spaced by means of feeder
spreaders. The feeder again is made of 300 ohm Twin -Lead since the feed -point impedance is approximately 300 ohms, the same as
that of the Twin -Lead folded dipole.
The folded -dipole type of antenna has the
broadest response characteristic (greatest
bandwidth) of any of the conventional halfwave antenna systems constructed of small
wires or conductors. Hence such an antenna
may be operated over the greatest frequency
range without serious standing waves of any
common half -wave antenna type.
The increased bandwidth of the multi -wire
doublet type of radiator, and the fact that the
feed -point resistance is increased sever al
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THE
Antennas and Antenna Matching
424
Z
e
STUB -FED
END -FED HERTZ
EPP
RADIO
O
A
-
300 -600
LINE
11
END -FED TYPES
r-O
0.95
A/2-1
--0.95 A/2--of-
TUNED DOUBLET
HALF-WAVE
300-600 O11M
STUB- FED
LINE
1
OPEN
QUARTER-WAVE
STUB- FED
SHORTED
300-60011 LINE
-0 .95
O
A/2
r--
-+{
O
T
FOUR -WIRE
0-FEO
0.95 9/2
0.95 9/2
--{
15011 TWINLEAD
0.193 OF FREE
TWIN LEAD
SPACE WAVELENGTH OR
FED
4
0.77 OF
LINE -FED
9/4
W---0.94
5/2{
3000
IN CENTER
O
2 -WIRE DOUBLET
DOUBLET
ANY LENGTH
ANY LENGTH
-0.95A/2--
FOR DELTA
f~--0.95 A/2
DIMENSIONS
SEE CHAP 19
DELTA MATCHED
300 OHM TWINLEAD
300 OHM TWINLEAD
-+I
ti
2 -OR
FEEDER
SPREADERS
OR `FOLDED DIPOLE
OLOEDDIPOIE
-
-0.95 5/2
6
TWINLEAD
LOW SIDE OPENED
n TWINLEAD
ANY LENGTH
0.94 5/2
Y1/4
O
TWINLEAD
300
60011 LINE
6000 LINE
-41
600 OHM LINE
ANY LENGTH
r--095
A/2--.{
14%
OF
0
TOTAL LENGTH
CO
STANDARD
-AA FED
DOUBLET
750
Figure 2
ALTERNATIVE
L
N 14 WIRE
TWINLEAD
ANY LENGTH
CENTER -FED TYPES,
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METHODS OF
FEEDING A
HALF -WAVE DIPOLE
Center -Fed Antennas
TI
ANT NUMBER OF HALF -WAVES
FROM TRANSMITTER
r
ANY EVEN NUMBER OF QUARTER -WAVES
f
TV
V
I
NIGH
CAPACITANCE /1
T
T LOW
CAPACITANCE
423
retuning the feeders. The overall efficiency of
the zepp antenna system is not quite as high
for long feeder lengths as for some of the antenna systems which employ non -resonant
transmission lines, but where space is limited
and where operation on more than one band
is desired, the zepp has some decided advantages.
As the radiating portion of the zepp antenna
system must always be some multiple of a
half wave long, there is always high voltage
present at the point where the live zepp feeder attaches to the end of the radiating portion
of the antenna. Thus, this type of zepp antenna system is voltage led.
I
Stub -Fed Zepp-
Figure
THE END -FED HERTZ ANTENNA
Showing the manner in which an end-fed Hertz
antenna may be fed through a low -impedance
line and low -pass filter by using a resonant
tank circuit as at (A), or through the use of
a reverse- connected pi network as at (B).
Type Radiator
1
Some harmonic -attenuating
provision (in addition to the usual low -pass TVI filter) must be
included in the coupling system, as an end fed antenna itself offers no discrimination
against harmonics, either odd or even.
The end -fed Hertz antenna has rather high
losses unless at least three -quarters of the
radiator can be placed outside the operating
room and in the clear. As there is r -f voltage
at the point where the antenna enters the
operating room, the insulation at that point
should be several times as effective as the
insulation commonly used with low- voltage
feeder systems. This antenna can be operated
on all of its higher harmonics with good efficiency, and can be operated at half frequency
against ground as a quarter -wave Marconi.
As the frequency of an antenna is raised
slightly when it is bent anywhere except at a
voltage or current loop, an end -fed Hertz antenna usually is a few per cent longer than a
straight half-wave doublet for the same frequency, because, ordinarily, it is impractical
to bring a wire in to the transmitter without
making several bends.
The zeppelin or zepp anterma system, illustrated
in figure 2A is very convenient when it is desired to operate a single
radiating wire on a number of harmonically reThe Zepp Antenna
System
lated frequencies.
The zepp antenna system is easy to tune,
and can be used on several bands by merely
a non -resonant
Figure 2C shows a modificaLion of the zepp -type antenna
system to allow the use of
transmission line between the
radiating portion of the antenna and the transmitter. The zepp portion of the antenna is
resonated as a quarter -wave stub and the non resonant feeders are connected to the stub at
a point where standing waves on the feeder
are minimized. The procedure for making these
adjustments is described in detail in Section
22 -8 This type of antenna system is quite
satisfactory when it is necessary physically
to end feed the antenna, but where it is necessary also to use non -resonant feeder between
the transmitter and the radiating system.
22 -2
Wave
Center -Fed Half Horizontal Antennas
The center feeding of a half -wave antenna
system is usually to be desired over an end fed system since the center-fed system is inherently balanced to ground and is therefore
less likely to be troubled by feeder radiation.
A number of center -fed systems are illustrated
in figure 2.
The Tuned
The current -fed do u b l e t with
spaced feeders, sometimes
called a center -fed zepp, is an
inherently balanced system if the two legs of
the radiator are electrically equal. This fact
holds true regardless of the frequency, or of
the harmonic, on which the system is operated. The system can successfully be operated over a wide range of frequencies if the
system as a whole (both tuned feeders and the
center -fed flat top) can be resonated to the
operating frequency. It is usually possible to
tune such an antenna system to resonance
with the aid of a tapped coil and a tuning caDoublet
pacitor that can optionally be placed either
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CHAPTER TWENTY -TWO
Antennas and Antenna Matching
Antennas for the lower frequency portion of
the h -f spectrum (perhaps from 1.8 to 7.0 Mc.),
and temporary or limited use antennas for the
upper portion of the h -f range, usually are of
a relatively simple type in which directivity
is not a prime consideration. Also, it often is
desirable, in amateur work, that a single antenna system be capable of operation at least
on the 3.5 -Mc. and 7.0 -Mc. range, and preferably on other frequency ranges. Consequently,
the first portion of this chapter will be devoted to a discussion of such antenna systems. The latter portion of the chapter is devoted to the general problem of matching the
antenna transmission line to antenna systems
of the fixed type. Matching the antenna transmission line to the rotatable directive array
is discussed in Chapter Twenty -five.
22 -1
End -Fed Half -Wave
Horizontal Antennas
Usually a high- frequency doublet is mounted
as high and as much in the clear as possible,
for obvious reasons. However, it is sometimes
justifiable to bring part of the radiating system directly to the transmitter, feeding the antenna without benefit of a transmission line.
This is permissible when (1) there is insufficient room to erect a 75- or 80 -meter horizontal dipole and feed line, (2) when a long wire
is also to be operated on one of the higher
frequency bands on a harmonic. In either case,
it is usually possible to get the main portion
of the antenna in the clear because of its
length. This means that the power lost by
bringing the antenna directly to the transmitter
is relatively small.
Even so, it is not best practice to bring the
high -voltage end of an antenna into the operating room because of the increased difficulty
in eliminating BC! and TVI. For this reason
one should dispense with a feed line in conjunction with a Hertz antenna only as a last
resort.
The end -fed antenna has no form
of transmission line to couple it
to the transmitter, but brings the
radiating portion of the antenna right down to
the transmitter, where some form of coupling
system is used to transfer energy to the anEnd -Fed
The half -wave horizontal dipole is the most
common and the most practical antenna for the
3.5 -Mc. and 7 -Mc. amateur bands. The form of
the dipole, and the manner in which it is fed
are capable of a large number of variations.
Figure 2 shows a number of practicable forms
of the simple dipole antenna along with methods of feed.
Antennas
tenna.
Figure 1 shows two common methods of
feeding the Fuchs antenna or end -fed Hertz.
422
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HANDBOOK
Tuned
amplitude, in turn, depends upon the mismatch
at the line termination. A line of no. 12 wire,
spaced 6 inches with good ceramic or plastic
spreaders, has a surge impedance of approximately 600 ohms, and makes an excellent
tuned feeder for feeding anything between 60
and 6000 ohms (at frequencies below 30 Mc.).
If used to feed a load of higher or lower impedance than this, the standing waves become
great enough in amplitude that some loss will
occur unless the feeder is kept short. At frequencies above 30 Mc., the spacing becomes
an appreciable fraction of a wavelength, and
radiation from the line no longer is negligible.
Hence, coaxial line or close- spaced parallel wire line is recommended for v -h -f work.
If a transmission line is not perfectly matched, it should be made resonant, even though
the amplitude of the standing waves (voltage
variation) is not particularly great. This prevents reactance from being coupled into the
final amplifier. A feed system having moderate
standing waves may be made to present a non reactive load to the amplifier either by tuning
or by pruning the feeders to approximate resonance.
Usually it is preferable with tuned feeders
to have a current loop (voltage minimum) at the
transmitter end of the line. This means that
when voltage- feeding an antenna, the tuned
feeders should be made an odd number of quarter wavelengths long, and when current -feeding
an antenna, the feeders should be made an
even number of quarter wavelengths long. Actually, the feeders are made about 10 per cent of
a quarter wave longer than the calculated
value (the value given in the tables) when
they are to be series tuned to resonance by
means of a capacitor, instead of being trimmed
and pruned to resonance.
When tuned feeders are used to feed an antenna on more than one band, it is necessary
to compromise and make provision for both
series and parallel tuning, inasmuch as it is
impossible to cut a feeder to a length that
will be optimum for several bands. If a voltage
loop appears at the transmitter end of the line
on certain bands, parallel tuning of the feeders will be required in order to get a transfer
of energy. It is impossible to transfer energy
by
inductive coupling unless current is flow-
ing. This is effected at a voltage loop by the
Lines
421
presence of the resonant tank circuit formed
by parallel tuning of the antenna' coil.
21
-12
Line Discontinuities
In the previous discussion we have assumed
transmission line which was uniform throughout its length. In actual practice, this is
usually not the case.
Whenever there is any sudden change in the
characteristic impedance of the line, partial
reflection will occur at the point of discontinuity. Some of the energy will be transmitted
and some reflected, which is essentially the
same as having some of the energy absorbed
and some reflected in so far as the effect upon
the line from the generator to that point is
concerned. The discontinuity can by ascribed
a reflection coefficient just as in the case of
an unmatched load.
In a simple case, such as a finite length of
uniform line having a characteristic impedance
of 500 ohms feeding into an infinite length of
uniform line having a characteristic impedance
of 100 ohms, the behavior is easily predicted.
The infinite 100 ohm lin& will have no standing
waves and will accept the same power from the
500 ohm line as would a 100 ohm resistor,
and the rest of the energy will be reflected at
the discontinuity to produce standing waves
from there back to the generator. However, in
the case of a complex discontinuity placed at
an odd distance down a line terminated in a
complex impedance, the picture becomes complicated, especially when the discontinuity is
neither sudden nor gradual, but intermediate
between the two. This is the usual case with
amateur lines that must be erected around
buildings and trees.
In any case, when a discontinuity exists
somewhere on a line and is not a smooth,
gradual change embracing several wavelengths,
it is not possible to avoid standing waves
throughout the entire length of the line. If the
discontinuity is sharp enough and is great
enough to be significant, standing waves must
exist on one side of the discontinuity, and
may exist on both sides in many cases.
a
www.americanradiohistory.com
420
line fed by
a
transmitter. It is the reflection
from the antenna end which starts waves moving back toward the transmitter end. When
waves moving in both directions along a conductor meet, standing waves are set up.
well- constructed open wire line has acceptably
Parallel -Wire Lines
low losses when its length
is less than about two wavelengths even when
the voltage standing -wave ratio is as high as
10 to 1. A transmission line constructed of
ribbon or tubular line, however, should have
the standing -wave ratio kept down to not more
than about 3 to 1 both to reduce power loss
and because the energy dissipation on the line
will be localized, causing overheating of the
line at the points of maximum current.
Because moderate standing waves can be
tolerated on open -wire lines without much loss,
a standing -wave ratio of 2/1 or 3/1 is considered acceptable with this type of line, even
when used in an untuned system. Strictly
speaking, a line is untuned, or non -resonant,
only when it is perfectly flat, with a standing wave ratio of 1 (no standing waves). However,
some mismatch can be tolerated with open-wire
untuned lines, so long as the reactance is not
objectionable, or is eliminated by cutting the
line to approximately resonant length.
Semi -Resonant
21 -11
THE
Radiation, Propagation and Lines
t
Zo
1;)
1.0
o
A
Tuned or
Resonant Lines
If a transmission line is terminated in its
characteristic surge impedance, there will be
no reflection at the end of the line, and the
current and voltage distribution will be uniform along the line. If the end of the line is
either open- circuited or short -circuited, the
reflection at the end of the line will be 100
per cent, and standing waves of very great amplitude will appear on the line. There will still
be practically no radiation from the line if it is
closely spaced, but voltage nodes will be
found every half wavelength, the voltage loops
corresponding to current nodes (figure 23).
If the line is terminated in some value of
resistance other than the characteristic surge
impedance, there will be some reflection, the
amount being determined by the amount of mismatch. With reflection, there will be standing
waves (excursions of current and voltage)
along the line, though not to the same extent
as with an open- circuited or short- circuited
line. The current and voltage loops will occur
at the same points along the line as with the
open or short- circuited line, and as the terminating impedance is made to approach the
characteristic impedance of the line, the cur-
RADIO
swR t.o ZL.
SWR
=
1.5
Zo
ZL.
+.s on 0.e1
Zo
1.s
o
SWR
3.0 ZL
3.0
OR
0.31 ZO
SWRo ZLooRo
Figure 23
STANDING WAVES ON A TRANSMISSION LINE
As shown at (A), the voltage and current are
constant on a transmission line which is
terminated in its characteristic impedance,
assuming that losses are small enough so
that they may be neglected. (B) shows the
variation in current or in voltage on a line
terminated in a load with a reflection coefficient of 0.2 so that a standing wave ratio
of 1.5 to I is set up. At (C) the reflection
coefficient has been increased to 0.5, with
the formation of a 3 to 1 standing -wove ratio
on the line. At (D) the line has been terminated in a load which has a reflection coefficient of I.0 (short, open circuit, or a pure
reactance) so that all the energy is reflected
with the formation of an infinite standing wave ratio.
rent and voltage along the line will become
more uniform. The foregoing assumes, of
course, a purely resistive (non -reactive) load.
If the load is reactive, standing waves also
will be formed. But with a reactive load the
nodes will occur at different locations from
the node locations encountered with wrong-
value resistive termination.
A well built 500- to 600 -ohm transmission
line may be used as a resonant feeder for
lengths up to several hundred feet with very
low loss, so long as the amplitude of the
standing waves (ratio of maximum to minimum
voltage along the line) is not too great. The
www.americanradiohistory.com
THE
RADIO
Transmission
ribbon and tubular configuration, with characteristic impedance values from 75 to 300
ohms. Receiving types, and transmitting types
for power levels up to one kilowatt in the h -f
range, are listed with their pertinent characteristics, in the table of figure 21.
419
Lines
204
Zo7SSLOGp
170
COAXIAL OR
CONCENTRIC LINE
us
Coaxial Line
Several types of coaxial cable
have come into wide use for
feeding power to an antenna system. A cross sectional view of a coaxial cable (sometimes
called concentric cable or line) is shown in
figure 22.
As in the parallel -wire line, the power lost
in a properly terminated coaxial line is the
sum of the effective resistance losses along
the length of the cable and the dielectric
losses between the two conductors.
Of the two losses, the effective resistance
loss is the greater; since it is largely due to
the skin effect, the line loss (all other conditions the same) will increase directly as the
square root of the frequency.
Figure 22 shows that, instead of having two
conductors running side by side, one of the
conductors is placed inside of the other. Since
the outside conductor completely shields the
inner one, no radiation takes place. The conductors may both be tubes, one within the
other; the line may consist of a solid wire
within a tube, or it may consist of a stranded
or solid inner conductor with the outer conductor made up of one or two wraps of copper
shielding braid.
In the type of cable most popular for military and non -commercial use the inner conductor consists of a heavy stranded wire, the
outer conductor consists of a braid of copper
wire, and the inner conductor is supported
within the outer by means of a semi -solid
dielectric of exceedingly low loss characteristics called polyethylene. The Army -Navy
designation on one size of this cable suitable
for power levels up to one kilowatt at frequencies as high as 30 Mc. is AN /RG -8 /U.
The outside diameter of this type of cable is
approximately one -half inch. The characteristic impedance of this cable type is 52 ohms,
but other similar types of greater and smaller
power-handling capacity are available in impedances of 52, 75, and 95 ohms.
When using solid dielectric coaxial cable
it is necessary that precautions be taken to
insure that moisture cannot enter the line. If
the better grade of connectors manufactured
for the line are employed as terminations, this
condition is automatically satisfied. If connectors are not used, it is necessary that
some type of moisture-proof sealing compound
be applied to the end of the cable where it
will be exposed to the weather.
Nearby metallic objects cause no loss, and
coaxial cable may be run up air ducts or ele-
loo
Di. INSIDE
70
DIAMETER OF
OUTER CONDUCTOR
s2
D= OUTSIDE DIAMETER OF
INNER CONDUCTOR
30
o
2.81
3.21
5
7
to
iS
30
RATIO OF DIAMETERS
Figure 22
CHARACTERISTIC IMPEDANCE OF AIR FILLED COAXIAL LINES
If the filling of the line is o dielectric material other than air, the characteristic impedance of the line will be reduced by a
factor proportional to the square -root of the
dielectric constant of the material used as a
dielectric within the line.
vator shafts, inside walls, or through metal
conduit. Insulation troubles can be forgotten.
The coaxial cable may be buried in the ground
or suspended above ground.
Standing Waves
Standing waves on a transmission line always are the
result of the reflection of energy. The only
significant reflection which takes place in a
normal installation is that at the load end of
the line. But reflection can take place from
discontinuities in the line, such as caused by
insulators, bends, or metallic objects adjacent
to an unshielded line.
When a uniform transmission line is terminated in an impedance equal to its surge impedance, reflection of energy does not occur,
and no standing waves are present. When the
load termination is exactly the same as the
line impedance, it simply means that the load
takes energy from the line just as fast as the
line delivers it, no slower and no faster.
Thus, for proper operation of an untuned
line (with standing waves eliminated), some
form of impedance- matching arrangement must
be used between the transmission line and
the antenna, so that the radiation resistance
of the antenna is reflected back into the line
as a nonreactive impedance equal to the line
impedance.
The termination at the antenna end is the
only critical characteristic about the untuned
www.americanradiohistory.com
418
RADIO
THE
Radiation, Propagation and Lines
CHARACTERISTICS OF COMMON TRANSMISSION LINES
ATTENUATION
db/
VELOCITYUUFD
00 FEET
vswR =1.0
1-
30 Mc 100 MC 300 MC
OPEN WIRE LINE,
COPPER.
N' 12
RIBBON LINE, REC.TYPE,
300 OHMS.
(7/2e
0.3
0.6
0.86
2.2
5.3
O.q8 -099
0.62
.,
1-271)
FT
Ny
6
RIBBON LINE, TRANS.
TYPE. 300 OHMS.
- - - - - - -
-
TUBULAR "TWIN -LEADTRANS. TYPE, 7/160.D. 0.65
(AMPHENOL 14 -076)
2 3
5.4
RIBBON LIKE, RECEIVE
TYPE, ISO OHMS.
2 7
6
1
1
O
REMARKS
PER
CONDUCTORS)
TUBULAR "TWIN-LEAD"
REC TYPE. 300 OHMS,
S /16.0.0., (AMPHENOL
TYPE
0.15
ACTOR
V
BASED UPON 4" SPACING BELOW 50 MC ; 2- SPACING ABOVE 50 MC. RADIATION
LOSSES INCLUDED. CLEAN, LOW LOSS CERAMIC INSULATION ASSUMED RADIATION
HIGH ABOVE 150 MC
FOR CLEAN. DRY LINE. wET WEATHER PERFORMANCE RATHER POOR BEST LINE IS
SLIGHTLY CONVEX. AVOID LINE THAT HAS CONCAVE DIELECTRIC SUITABLE FOR
LOW POWER TRANSMITTING APPLICATIONS. LOSSES INCREASE AS LINE WEATHERS.
HANDLES 400 WATTS AT 30 MC. IF VSWR IS LOW.
CHARACTERISTICS SIMILAR TO RECEIVING TYPE RIBBON LINE EXCEPT FOR MUCH
BETTER wET WEATHER PERFORMANCE.
CHARACTERISTICS VARY SOMEWHAT WITH MANUFACTURER. BUT APPROXIMATE
THOSE OF RECEIVING TYPE RIBBON EXCEPT FOR GREATER POWER HANDLING
CAPABILITY AND SLIGHTLY BETTER WET WEATHER PERFORMANCE.
0.79
8.1
FOR USE WHERE RECEIVING TYPE TUBULAR -TWIN -LEAD DOES NOT HAVE SUMCIENT POWER HANDLING CAPABILITY. WILL HANDLE / KW AT 30 MC. F VSWR
IS LOW.
0 77 V'
10
USEFUL FOR QUARTER WAVE MATCHING SECTIONS.
I
AS A
NO LONGER WIDELY USED
LINE.
USEFUL MAINLY IN THE H -F RANGE BECAUSE OF EXCESSIVE LOSSES AT V -H -F
AND U-H-F. LESS AFFECTED BY WEATHER THAN 300 OHM_RIBBON.
VERY SATISFACTORY FOR TRANSMITTING APPLICATIONS BELOW 30 MC. AT
KW. NOT SIGNIFICANTLY AFFECTED BY WET WEATHER.
POWERS UP TO
RIBBON LINE, RECEIVE.
TYPE, 75 OHMS.
2
5 O
11.0
o.BB'
19 V'
RIBBON LINE, TRANS.
TYPE, 75 OHMS.
1.5
3.9.
6.0
0.71Y
f
RG-6/U COAX (52
OHMS)
1.0
2.1
4.2
0.88
29.5
WILL HANDLE 2 KW AT 4O MC. IF VSWR
RG-11 /U COAX (75 OHMS)
0.94
I 9
3.6
0.88
20.5
WILL HANDLE
1.
RG -17 /U COAX (520HMS)
0.38
0.85
1.8
0.66
29.5
WILL HANDLE
7
1.95
4.1
8.0
0.66
28.5
WILL HANDLE 430 WATTS AT 30 MC. IF VSWR IS LOW. 0.2000.
RG
-58/U
COAX
(53 OHMS)
O
6`t
1
KW AT
IS LOW.
30 MC. IF VSWR
0..
IS LOW.
8 KW. AT 30 MC. IF VSWR IS LOW.
0
O.D.
7/21 CONDUCTOR.
4 "0.0. 7/28 CONDUCTOR.
087" OD. 0.19"
0.24" O.D.
DIA. CONDUCTOR
N 20 CONDUCTOR.
N 22 CONDUCTOR.
RG-S9 /U COAX (73 OHMS)
1.9
3.8
7.0
0.66
21
WILL HANDLE 680 WATTS AT 30 MC. IF VSWR
(720HMS)
2.0
4.0
7.0
0.66
22
COMMERCIAL VERSION OF RG-59/U FOR LESS EXACTING APPLICATIONS.
EXPENSIVE.
FOR SHIELDED, BALANCED -TO- GROUND APPLICATIONS. VERY LOW NOISE
PICK UP. 0.4" 0.D.
TV -59 COAX
RG -22/U
SHIELDED
PAIR (95 OHMS)
K
-I11 SHIELDED PAIR
(300 OHMS)
1.7
3.0
5.5
0.66
18
2.0
3.5
6.1
-
4
0
APPROXIMATE. EXACT FIGURE VARIES SLIGHTLY WITH MANUFACTURER
2S
= 276
1og10d
Where:
S is the exact distance between wire centers
in some convenient unit of measurement, and
d is the diameter of the wire measured in the
same units as the wire spacing, S.
2S
Since
LESS
DESIGNED FOR TV LEAD -IN IN NOISY LOCATIONS. LOSSES HIGHER THAN
REGULAR 300 OHM RIBBON, BUT DO NOT INCREASE AS MUCH FROM WEATHERING
FIGURE
Z.
IS LOW.
21
Surge impedance values of less than 200
ohms are seldom used in the open -type two wire line, and, even at this rather high value
of Z. the wire spacing S is uncomfortably
close, being only 2.7 times the wire diameter d.
Figure 20 gives in graphical form the surge
impedance of practicable two -wire lines. The
chart is self-explanatory, and is sufficiently
accurate for practical purposes.
Instead of using spacer insulators placed periodically
along the transmission line
it is possible to mold the
line conductors into a ribbon or tube of flexible low -loss dielectric material. Such line,
with polyethylene dielectric, is used in enormous quantities as the lead -in transmission
line for FM and TV receivers. The line is
available from several manufacturers in the
Ribbon and
expresses
a
ratio only, the units
d
of measurement may be centimeters, millimeters, or inches. This makes no difference
in the answer, so long as the substituted
values for S and d are in the same units.
The equation is accurate so long as the
wire spacing is relatively large as compared
to the wire diameter.
Tubular Transmission Line
www.americanradiohistory.com
HANDBOOK
Transmission
ever, mechanical or electrical considerations
often make one type of transmission line better
adapted for use to feed a particular type of
antenna than any other type.
Transmission lines for carrying r -f energy
are of two general types: non -resonant and
resonant. A non -resonant transmission line
is one on which a successful effort has been
made to eliminate reflections from the termination (the antenna in the transmitting case
and the receiver for a receiving antenna) and
hence one on which standing waves do not
exist or are relatively small in magnitude. A
resonant line, on the other hand, is a transmission line on which standing waves of appreciable magnitude do appear, either through
inability to match the characteristic impedance
of the line to the termination or through intentional design.
The principal types of transmission line in
use or available at this time include the open wire line (two -wire and four -wire types), two wire solid -dielectric line ( "Twin- Lead" and
similar ribbon or tubular types), two -wire polyethylene- filled shielded line, coaxial line of
the solid -dielectric, beaded, stub-supported,
or pressurized type, rectangular and cylindrical
wave guide, and the single -wire feeder operated against ground. The significant characteristics of the more popular types of transmission line available at this time are given
in the chart of figure 21.
21 -10
Non -Resonant
Transmission Lines
A non -resonant or untuned transmission
line
is a line with negligible standing waves.
Hence, a non -resonant line is a line carrying
r -f power only in one direction -from the source
of energy to the load.
Physically, the line itself should be identical throughout its length. There will be a
smooth distribution of voltage and current
throughout its length, both tapering off very
slightly towards the load end of the line as a
result of line losses. The attenuation (loss)
in certain types of untuned lines can be kept
very low for line lengths up to several thousand feet. In other types, particularly where
the dielectric is not air (such as in the twisted pair line), the losses may become excessive
at the higher frequencies, unless the line is
relatively short.
Transmission -Line
All transmission lines have
distributed
inductance,
capacitance and resist ance. Neglecting the resistance, as it is of
minor importance in short lines, it is found
Impedance
Lines
417
1111111111ESSMINI%s's_í.s
%%M III /
IIM/11111Ï11
ILW
/Oií%E'/_Er1111111111
.--
iiilMEM ;mt=1
s
.
io
is
3
s
INCHES. CENTER TO CENTER
111111111
+
w
w
u
Figure 20
CHARACTERISTIC IMPEDANCE OF TYPICAL TWO -WIRE OPEN LINES
that the inductance and capacitance per unit
length determine the characteristic or surge
impedance of the line. Thus, the surge impedance depends upon the nature and spacing
of the conductors, and the dielectric separating them.
Speaking in electrical terms, the characteristic impedance of a transmission line is
simply the ratio of the voltage across the line
to the current which is flowing, the same as
is the case with a simple resistor: Z. = E /1.
Also, in a substantially loss -less line (one
whose attenuation per wavelength is small)
the energy stored in the line will be equally
divided between the capacitive field and the
inductive field which serve to propagate the
energy along the line. Hence the characteristic impedance of a line maybe expressed as:
Z.
Two -Wire
Open Line
=
N/
L/C.
two -wire transmission system
is easy to construct. Its surge impedance can be calculated quite
easily, and when properly adjusted and balanced to ground, with a conductor spacing
which is negligible in terms of the wavelength of the signal carried, undesirable feeder
radiation is minimized; the current flow in
the adjacent wires is in opposite directions,
and the magnetic fields of the two wires are
in opposition to each other. When a two -wire
line is terminated with the equivalent of a
pure resistance equal to the characteristic
impedance of the line, the line becomes a non resonant line.
Expressed in physical terms, the characteristic impedance of a two -wire open line is
A
equal to:
www.americanradiohistory.com
416
THE
Radiation, Propagation and Lines
dent, particularly a "flutter fade" and a characteristic "hollow" or echo effect.
Deviations from a great circle path are especially noticeable in the case of great circle
paths which cross or pass near the auroral
zones, because in such cases there often is
complete or nearly complete absorption of the
direct sky wave, leaving off -path scattered
reflections the only mechanism of propagation.
Under such conditions the predominant wave
will appear to arrive from a direction closer
to the equator, and the signal will be noticeably if not considerably weaker than a direct
sky wave which is received under favorable
conditions.
Irregular reflection of radio waves from
"scattering patches" is divided into two categories: "short scatter" and "long scatter ".
Short scatter is the scattering that occurs
when a radio wave first reaches the scattering
patches or media. Ordinarily it is of no particular benefit, as in most cases it only serves
to fill in the inner portion of the skip zone
with a weak, distorted signal.
Long scatter occurs when a wave has been
refracted from the F2 layer and strikes scattering patches or media on the way down. When
the skip distance exceeds several hundred
miles, long scatter is primarily responsible
for reception within the skip zone, particularly the outer portion of the skip zone. Distortion is much less severe than in the case
of short scatter, and while the signal is likewise weak, i t sometimes can be utilized for
satisfactory communication.
During a severe ionosphere disturbance in
the north auroral zone, it sometimes is possible
to maintain communication between the Eastern
United States and Northern Europe by the following mechanism: That portion of the energy
which is radiated in the direction of the great
circle path is completely absorbed upon reaching the auroral zone. However, the portion of
the wave leaving the United States in a southeasterly direction is refracted downward from
the F2 layer and encounters scattering patches
or media on its downward trip at a distance
of approximately 2000 miles from the transmitter. There it is reflected by "long scatter"
in all directions, this scattering region acting
like an isotropic radiator fed with a very small
fraction of the original transmitter power. The
great circle path from this southerly point to
northern Europe does not encounter unfavorable ionosphere conditions, and the wave is
propagated the rest of the trip as though it had
been radiated from the scattering region.
Another type of scatter is produced when
a sky wave strikes certain areas of the earth.
Upon striking a comparatively smooth surface
such as the sea, there is little scattering, the
wave being shot up again by what could be
RADIO
considered specular or mirror reflection. But
upon striking a mountain range, for instance,
the reradiation or reflected energy is scattered,
some of it being directed back towards the
transmitter, thus providing another mechanism
for producing a signal within the skip zone.
strikes the earth's
atmosphere, a cylindrical region
of free electrons is formed at
approximately the height of the E layer. This
slender ionized column is quite long, and when
first formed is sufficiently dense to reflect
radio waves back to earth most readily, including v -h -f waves which are not ordinarily
returned by the F= layer.
The effect of a single meteor, of normal size,
shows up as a sudden "burst" of signal of
short duration at points not ordinarily reached
by the transmitter. After a period of from 10
to 40 seconds, recombination and diffusion
have progressed to the point where the effect
of a single fairly large meteor is not perceptible. However, there are many small meteors
impinging upon earth's atmosphere every minute, and the aggregate effect of their transient
ionized trails, including the small amount of
residual ionization that exists for several
minutes after the original flash but is too weak
and dispersed to prolong a "burst ", is believed to contribute to the existence of the
"nighttime E" layer, and perhaps also to
sporadic E patches.
While there are many of these very small
meteors striking the earth's atmosphere every
minute, meteors of normal size (sufficiently
large to produce individual "bursts ") do not
strike nearly so frequently except during some
of the comparatively rare meteor "showers ".
Metéors and
When a meteor
"Bursts"
During one of these displays a "quivering"
ionized layer is produced which is intense
enough to return signals in the lower v -h -f
range with good strength, but with a type of
"flutter" distortion which is characteristic
of this type of propagation.
21 -9
Transmission Lines
For many reasons it is desirable to place
an antenna or radiating system as high and in
the clear as is physically possible, utilizing
some form of nonradiating transmission line
to carry energy with as little loss as possible
from the transmitter to the radiating antenna,
and conversely from the antenna to the re-
ceiver.
There are many different types of transmission lines and, generally speaking, practically
any type of transmission line or feeder system
may be used with any type of antenna. How-
HANDBOOK
11 -Year
Sunspot Cycle
415
225
200
i'
175
150
é
125
50
'5
ó;
C
;
I
,
2
,-`
,
.
3
25
YEAR
o
48
ttt
ri
100
75
,
50
52
54
58
Figure
56
80
62
64
86
18
THE YEARLY TREND OF THE SUNSPOT
CYCLE. RADIO CONDITIONS IN GENERAL WILL DETERIORATE DURING 19601965 AS THE CYCLE DECLINES.
zon, the farther away will the wave return to
earth, and the greater the skip distance. The
wave can be reflected back up into the ionosphere by the earth, and then be reflected back
down again, causing a second skip distance
area. The drawing of figure 19 shows the multiple reflections possible. When the receiver
receives signals which have traveled over
more than one path between transmitter and
receiver, the signal impulses will not all arrive
at the same instant, as they do not all travel
the same distance. When two or more signals
arrive in the same phase at the receiving antenna, the resulting signal in the receiver will
be quite strong. On the other hand, if the signals arrive 180° out of phase, so they tend to
cancel each other, the received signal will
drop -perhaps to zero if perfect cancellation
occurs. This explains why high -frequency
signals are subject to fading.
Fading can be greatly reduced on the high
frequencies by using a transmitting antenna
with sharp vertical directivity, thus cutting
down the number of possible paths of signal
arrival. A receiving antenna with similar characteristics (sharp vertical directivity) will
further reduce fading. It is desirable, when
using antennas with sharp vertical directivity,
to use the lowest vertical angle consistent
with good signal strength for the frequency
used.
Scattered
Reflections
Scattered reflections are random,
diffused, substantially isotropic
reflections which are partly re-
TRANSMITTER
Figure 19
IONOSPHERE -REFLECTION WAVE PATHS
Showing typical ionosphere- reflection wave
paths during daylight hours when ionization
density is such that frequencies as high as
28 Mc. will he returned to earth. The distance between ground -wave range and that
range where the ionosphere -reflected wove of
a specific frequency first will be returned to
earth is called the skip distance.
sponsible for reception within the skip zone,
and for reception of signals from directions
off the great circle path.
In a heavy fog or mist, it is difficult to see
the road at night because of the bright glare
caused by scattered reflection of the headlight beam by the minute droplets. In fact, the
road directly to the side of the car will be
weakly illuminated under these conditions,
whereas it would riot on a clear night (assuming flat, open country). This is a good example
of propagation of waves by scattered reflections into a zone which otherwise would not
be illuminated.
Scattering occurs in the ionosphere at all
times, because of irregularities in the medium (which result in "patches" corresponding
to the water droplets) and because of random phase radiation due to the collision or recombination of free electrons. However, the nature
of the scattering varies widely with time, in
a random fashion. Scattering is particularly
prevalent in the f: region, but scattered reflections may occur at any height, even well
out beyond the virtual height of the /-'2 layer.
There is no "critical frequency" or "lowest perforating frequency" involved in the
scattering mechanism, though the intensity
of the scattered reflections due to typical
scattering in the F. region of the ionosphere
decreases with frequency.
CChen the received signal is due primarily
to scattered reflections, as is the case in the
skip zone or where the great circle path does
not provide a direct sky wave (due to low critical or perforation frequency, or to an ionosphere storm) very bad distortion will be evi-
www.americanradiohistory.com
10
34
32
30
WINTER
2e
SUNSPOT
MAXIMUM
26
l'
1
24
22
ï
20
U
16
w
14
a
12
ù
Z
rc
a
T
Radiation, Propagation and Lines
414
-.1,,
III
SUMMER
SUNSPOTMINIMUM
-
4
2
0
2
6
e
10
12
14
16
16
ii'
20
22
24
LOCAL TIME
Figure
17
TYPICAL CURVES SHOWING CHANGE
M.U.F.
IN
AT
MINIMUM
MAXIMUM
AND
POINTS IN SUNSPOT CYCLE
The m.u.f. often drops to frequencies below
the early morning hours. The high
m.u.f. in the middle of the day is brought about
by reflection from the F2 layer. M.u.f. data is
published periodically in the magazines devoted to amateur work, and the m.u.f. can be
calculated with the aid of Basic Radio Propagation Predictions, CRPL -D, published monthly by the Government Printing Office, Washington, D.C.
10 Mc. in
The optimum working frequency for any particular
Frequency
direction and distance is
usually about 15 per cent
less than the m.u.f. for contact with that particular location. The absorption by the ionosphere becomes greater and greater as the
operating frequency is progressively lowered
below the m.u.f. It is this condition which
causes signals to increase tremendously in
strength on the 14 -Mc. and 28 -Mc. bands just
before the signals drop completely out. At the
time when the signals are greatest in amplitude the operating frequency is equal to the
m.u.f. Then as the signals drop out the m.u.f.
has become lower than the operating frequency.
Absorption and
Optimum Working
The shortest distance from a
transmitting location at which
signals reflected from the ionosphere can be
returned to the earth is called the skip distance. As was mentioned above under Critical
Frequency there is no skip distance for a frequency below the critical frequency of the
Skip Distance
E
R A D
I
O
most highly ionized layer of the ionosphere
at the time of transmission. However, the skip
distance is always present on the 14 -Mc. band
and is almost always present on the 3.5 -Mc.
and 7 -Mc. bands at night. The actual measure
of the skip distance is the distance between
the point where the ground wave falls to zero
and the point where the sky wave begins to
return to earth. This distance may vary from
40 to 50 miles on the 3.5-Mc. band to thousands of miles on the 28-Mc. band.
Occasional patches of extremely high ionization density appear at intervals
throughout the year at a height approximately
equal to that of the F layer. These patches,
called the sporadic-F. layer may be very small
or may be up to several hundred miles in extent. The critical frequency of the sporadic-F
layer may be greater than twice that of the
normal ionosphere layers which exist at the
The Sporadic -E
Loyer
e
e
H
same time.
It is this sporadic -E condition which provides "short- skip" contacts from 400 to perhaps 1200 miles on the 28 -Mc. band in the
evening. It is also the sporadic-E condition
which provides the more common type of "band
opening" experienced on the 50 -Mc. band when
very loud signals are received from stations
from 400 to 1200 miles distant.
Cycles in
The ionization density of
the ionosphere is deterIonosphere Activity
mined by the amount of
radiation (probably ultra violet) which is being received from the sun. Consequently, ionosphere activity is a function of the amount of
radiation of the proper character being emitted
by the sun and is also a function of the relative aspect of the regions in the vicinity of
the location under discussion to the sun. There
are four main cycles in ionosphere activity.
These cycles are: the daily cycle which is
brought about by the rotation of the earth, the
27 -day cycle which is caused by the rotation
of the sun, the seasonal cycle which is caused
by the movement of the earth in its orbit, and
the 11-year cycle which is a cycle in sunspot
activity. The effects of these cycles are superimposed insofar as ionosphere activity is concerned. Also, the cycles are subject to short
term variations as a result of magnetic storms
and similar terrestrial disturbances.
The most recent minimum of the 11 -year sunspot cycle occurred during the winter of 19541955, and we are currently moving along the
slope of a new cycle, the maximum of which
occurred during 1958. The current cycle is pictured in figure 18.
Fading
The lower the angle of radiation of
the wave, with respect to the hori-
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HANDBOOK
Ionospheric Propagation
200
virtual height of approximately 175 miles at
night, and in the daytime it splits up into two
layers, the upper one being called the F, layer
and the lower being called the F, layer. The
height of the F2 layer during daylight hours is
normally about 250 miles on the average and
the F, layer often has a height of as low as
140 miles. It is the F2 layer which supports
all nighttime dx communication and nearly all
daytime dx propagation.
a
F2
130
Ft
too
MID DAY
E
D
F2
Below the F2 layer is another
layer, called the E layer, which
is of importance in daytime communication
over moderate distances in the frequency range
between 3 and 8 Mc. This layer has an almost
constant height at about 70 miles. Since the
re- combination time of the ions at this height
is rather short, the E layer disappears almost
completely a short time after local sunset.
The
MIDNIGHT
too
50
IONIZATION DENSITY
-a
Figure 16
IONIZATION DENSITY IN THE IONOSPHERE
Showing typical ionization density of the
ionosphere in mid -summer. Note that the Ft
and D layers disappear at night, and that the
density of the E layer falls to such a low
value that it is ineffective.
which the sky wave can undergo depends upits frequency, and the amount of ionization
in the ionosphere, which is in turn dependent
upon radiation from the sun. The sun increases
the density of the ionosphere layers (figure 16)
and lowers their effective height. For this
reason, the ionosphere acts very differently
at different times of day, and at different times
of the year.
The higher the frequency of a radio wave,
the farther it penetrates the ionosphere, and
the less it tends to be bent back toward the
earth. The lower the frequency, the more easily
the waves are bent, and the less they penetrate the ionosphere. 160 -meter and 80-meter
signals will usually be bent back to earth
even when sent straight up, and may be considered as being reflected rather than refracted. As the frequency is raised beyond about
5,000 kc. (dependent upon the critical frequency of the ionosphere at the moment), it is
found that waves transmitted at angles higher
than a certain critical angle never return to
earth. Thus, on the higher frequencies, it is
necessary to confine radiation to low angles,
since the high angle waves simply penetrate
the ionosphere and are lost.
on
The F2 Layer
413
The
higher of the two major
reflection regions of the ionosphere is called the F, layer. This layer has
E
Layer
Below the E layer at a height of
about 35 miles is an absorbing
layer, called the D layer, which exists in the
middle of the day in the summertime. The layer
also exists during midday in the winter time
during periods of high solar activity, but the
layer disappears completely at night. It is this
layer which causes high absorption of signals
in the medium and high- frequency range during
the middle of the day.
The
D
Layer
Critical Frequency
critical frequency of
ionospheric layer is the
highest frequency which will be reflected when
the wave strikes the layer at vertical incidence. The critical frequency of the most highly ionized layer of the ionosphere may be as
low as 2 Mc. at night and as high as 12 to 13
Mc. in the middle of the day. The critical frequency is directly of interest in that a skip distance zone will exist on all frequencies
greater than the highest critical frequency at
that time. The critical frequency is a measure
of the density of ionization of the reflecting
layers. The higher the critical frequency the
greater the density of ionization.
The
an
Maximum Usable
The maximum usable /requency or m.u. /. is of great
importance in long- distance
communication since this frequency is the highest that can be used for communication between any two specified areas. The m.u.f. is
the highest frequency at which a wave projected into space in a certain direction will
be returned to earth in a specified region by
ionospheric reflection. The m.u.f. is highest
at noon or in the early afternoon and is highest in periods of greatest sunspot activity,
often going to frequencies higher than 50 Mc.
Frequency
(figure 17).
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THE
Radiation, Propagation and Lines
412
T
__
INVERSION
DUCT
T
INVERSION AND DUCT
REFRACTIVE INDEX
Figure
15
ILLUSTRATING DUCT TYPES
Showing two types of variation in refractive
index with height which will give rise to the
formation of a duct. An elevated duct is
shown at (A), and o ground -based duct is
shown at (B). Such ducts can propagate
ground-wave signals far beyond their normal
range.
rise to the formation of a duct which
can propagate waves with very little attenuation over great distances in a manner similar
to the propagation of waves through a wave
guide. Guided propagation through a duct in
the atmosphere can give quite remarkable
transmission conditions (figure 15). However,
such ducts usually are formed only on an over water path. The depth of the duct over the
water 's surface may be only 20 to 50 feet, or
it may be 1000 feet deep or more. Ducts exhibit a low -frequency cutoff characteristic
similar to a wave guide. The cutoff frequency
is determined by depth of the duct and by the
strength of the discontinuity in refractive index at the upper surface of the duct. The low est'frequency that can be propagated by such
a duct seldom goes below 50 Mc., and usually
will be greater than 100 Mc. even along the
may give
Pacific Coast.
virtue of
Stratospheric
Communication
Reflection
stratospheric reflection can be
by
brought about during magnetic
storms, aurora borealis displays, and during
meteor showers. Dx communication during extensive meteor showers is characterized by
frequent bursts of great signal strength followed by a rapid decline in strength of the
received signal. The motion of the meteor
forms an ionized trail of considerable extent
which can bring about effective reflection of
signals. However, the ionized region persists
only for a matter of seconds so that a shower
of meteors is necessary before communication
becomes possible.
The type of communication which is possible
during visible displays of the aurora borealis
RADIO
and during magnetic storms has been called
aurora-type dx. These conditions reach a maximum somewhat after the sunspot cycle peak,
possibly because the spots on the sun are
nearer to its equator (and more directly in line
with the earth) in the latter part of the cycle.
Ionospheric storms generally accompany magnetic storms. The normal layers of the ionosphere may be churned or broken up, making
radio transmission over long distances difficult or impossible on high frequencies. Unusual conditions in the ionosphere sometimes
modulate v -h-f waves so that a definite tone or
noise modulation is noticed even on transmitters located only a few miles away.
A pecularity of this type of auroral propagation of v -h -f signals in the northern hemisphere
is that directional antennas usually must be
pointed in a northerly direction for best results
for transmission or reception, regardless of
the direction of the other station being contacted. Distances out to 700 or 800 miles have
been covered during magnetic storms, using
30 and 50 Mc. transmitters, with little evidence of any silent zone between the stations
communicating with each other. Generally,
voice -modulated transmissions are difficult or
impossible due to the tone or noise modulation
on the signal. Most of the communication of
this type has taken place by c.w. or by tone
modulated waves with a keyed carrier.
21-8
Ionospheric
Propagation
Propagation of radio waves for communication on frequencies between perhaps 3 and
30 Mc. is normally carried out by virtue of
ionospheric reflection or refraction. Under conditions of abnormally high ionization in the
ionosphere, communication has been known
to have taken place by ionospheric reflection
on frequencies higher than 50 Mc.
The ionosphere consists of layers of ionized
gas located above the stratosphere, and extending up to possibly 300 miles above the
earth. Thus we see that high- frequency radio
waves may travel over short distances in a
direct line from the transmitter to the receiver,
or they can be radiated upward into the ionosphere to be bent downward in an indirect ray,
returning to earth at considerable distance
from the transmitter. The wave reaching a receiver via the ionosphere route is termed a
sky wave. The wave reaching a receiver by
traveling in a direct line from the transmitting
antenna to the receiving antenna is commonly
called a ground wave.
The amount of bending at the ionosphere
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HANDBOOK
TRANSMITTING
ANTENNA
Ground Wave
Di
DIRECT WAVES
e
e
GROUND- REFLECTED
WAVES
D2
D3
RECEIVING
ANTENNA
AT DIFFERENT
HEIGHTS
Figure 14
INTERFERENCE WITH HEIGHT
When the source of a horizontally -polarized
space -wave signal is above the horizon, the
received signal at a distant location will go
through a cyclic variation as the antenna
height is progressively raised. This is due
to the difference in total path length between
the direct wove and the ground-reflected
wave, and to the fact that this path length
difference changes with antenna height.
When the path length difference is such that
the two waves arrive at the receiving antenna with a phase difference of 3600 or some
multiple of 3600, the two waves will appear
WAVE
to be in phase as for as the antenna is concerned and maximum signal will be obtained.
On the other hand, when the antenna height
is such that the path length difference for
the two waves causes the waves to arrive
with a phase difference of an odd multiple
of 1800 the two waves will substantially cancel, and a null will be obtained at that antenna height. The difference between DI
and D2 plus D3 is the path- length difference.
Note also that there is an additional 1800
phase shift in the ground-reflected wave at
the point where it is reflected from the
ground. It is this latter phase shift which
causes the space-wave field intensity of a
horizontally polarized wave to be zero with
the receiving antenna at ground level.
is in miles and the antenna height N is in
feet. This equation must be applied separately
to the transmitting and receiving antennas and
the results added. However, refraction and
diffraction of the signal around the spherical
earth cause a smaller reduction in field strength
than would occur in the absence of such bending, so that the average radio horizon is somewhat beyond the geometrical horizon. The
equation d = 1.4 N,/ f is sometimes used for
determining the radio horizon.
d
Tropospheric
Propagation
Propagation by signal bending
in the lower atmosphere, called
tropospheric propagation,
can
result in the reception of signals over a much
greater distance than would be the case if the
lower atmosphere were homogeneous. In a
homogeneous or well -mixed lower atmosphere,
called a normal or standard atmosphere, there
is a gradual and uniform decrease in index of
refraction with height. This effect is due to
Communication
411
the combined effects of a decrease in temperature, pressure, and water -vapor content with
height.
This gradual decrease in refractive index
with height causes waves radiated at very low
angles with respect to the horizontal to be
bent downward slightly in a curved path. The
result of this effect is that such waves will be
propagated beyond the true or geometrical
horizon. In a so- called standard atmosphere
the effect of the curved path is the same as
though the radius of the earth were increased
by approximately one third. This condition extends the horizon by approximately 30 per cent
for normal propagation, and the extendedhorizon is known as the radio path horizon, mentioned before.
Conditions Leading to
Tropospheric
Stratification
When the temperature,
pressure, or water-vapor
content of the atmos-
phere does not change
smoothly with rising altitude, the discontinuity
or stratification will result in the reflection
or refraction of incident v -h -f signals. Ordinarily this condition is more prevalent at night
and in the summer. In certain areas, such as
along the west coast of North America, it is
frequent enough to be considered normal. Signal strength decreases slowly with distance
and, if the favorable condition in the lower
atmosphere covers sufficient area, the range
is limited only by the transmitter power, antenna gain, receiver sensitivity, and signal -tonoise ratio. There is no skip distance. Usually,
transmission due to this condition is accompanied by slow fading, although fading can be
violent at a point where direct waves of about
the same strength are also received.
Bending in the troposphere, which refers to
the region from the earth's surface up to about
10 kilometers, is more likely to occur on days
when there are stratus clouds than on clear,
cool days with a deep blue sky. The temperature or humidity discontinuities may be broken
up by vertical convection currents over land
in the daytime but are more likely to continue
during the day over water. This condition is
in some degree predictable from weather information several days in advance. It does not
depend on the sunspot cycle. Like direct communication, best results require similar antenna polarization or orientation at both the
transmitting and receiving ends, whereas in
transmission via reflection in the ionosphere
(that part of the atmosphere between about 50
and 500 kilometers high) it makes little difference whether antennas are similarly polarized.
Duct Formation
www.americanradiohistory.com
When bending conditions are
particularly
favorable
they
41
THE
Radiation, Propayation and Lines
0
RADIO
bination of the two. The three waves which
may combine to make up the ground wave are
illustrated in figure 13.
@DIRECT WAVE
©GROUND -REFLECTED
l--- ' - - -
WAVE
@SURFACE WAVE
Figure
-
13
GROUND -WAVE SIGNAL PROPAGATION
The
illustration above shows the three com-
ponents of the ground wave: (A), the surface
wave; (B), the direct wave; and (C), the
ground-reflected wove. The direct wave and
the
ground -reflected
receiving
antenna
wove combine at the
up the space
to make
wive.
may take place as a result of the ground wave,
or as a result of the sky wave or ionospheric
wave.
The term ground wave actually includes several different types of waves which usually are called:
(1) the surface wave, (2) the direct wave, and
(3) the ground -reflected wave. The latter two
waves combine at the receiving antenna to
form the resultant wave or the space wave.
The distinguishing characteristic of the components of the ground wave is that all travel
along or over the surface of the earth, so that
they are affected by the conductivity and terrain of the earth's surface.
The Ground Wave
Intense bombardment of
the upper regions of
the atmosphere by radiations from the sun results in the formation
of ionized layers. These ionized layers, which
form the ionosphere, have the capability of
reflecting or refracting radio waves which impinge upon them. A radio wave which has been
propagated as a result of one or more reflections from the ionosphere is known as an
ionospheric wave or a sky wave. Such waves
make possible long distance radio communication. Propagation of radio signals by ionospheric waves is discussed in detail in Sec-
The Ionospheric Wove
or Sky Wave
tion 21 -8.
21 -7
Ground -Wave
Communication
As stated in the preceding paragraph, the
term ground wave applies both to the surface
wave and to the space wave (the resultant
wave from the combination of the direct wave
and the ground -reflected wave) or to a com-
The Surface Wove
The surface wave is that
wave which we normally
receive from a standard broadcast station. It
travels directly along the ground and terminates on the earth's surface. Since the earth is
a relatively poor conductor, the surface wave
is attenuated quite rapidly. The surface wave
is attenuated less rapidly as it passes over
sea water, and the attenuation decreases for
a specific distance as the frequency is decreased. The rate of attenuation with distance
becomes so large as the frequency is increased
above about 3 Mc. that the surface wave becomes of little value for communication.
The resultant wave or space
wave is illustrated in figure
13 by the combination of (B) and (C). It is this
wave path, which consists of the combination
of the direct wave and the ground-reflected
wave at the receiving antenna, which is the
normal path of signal propagation for line -ofsight or near line -of -sight communication or
FM and TV reception on frequencies above
about 40 Mc.
Below line-of -sight over plane earth or
water, when the signal source is effectively
at the horizon, the ground-reflected wave does
not exist, so that the direct wave is the only
component which goes to make up the space
wave. But when both the signal source and the
receiving antenna are elevated with respect to
the intervening terrain, the ground-reflected
wave is present and adds vectorially to the
direct wave at the receiving antenna. The vectorial addition of the two waves, which travel
over different path lengths (since one of the
waves has been reflected from the ground) results in an interference pattern. The interference between the two waves brings about a
cyclic variation in signal strength as the receiving antenna is raised above the ground.
This effect is illustrated in figure 14. From
this figure it can be seen that best space wave reception of a v -h -f signal often will be
obtained with the receiving antenna quite close
to the ground. This subject, along with other
aspects of v -h -f signal propagation and reception, are discussed in considerable detail in
a book on fringe -area TV reception.
The distance from an elevated point to the
geometrical horizon is gitiren by the approximate equation: d = 1.221 where'the distance
The Space Wave
"Better TV Reception," by W. W. Smith and R. L. Dowley, published by Editors and Engineers, Ltd., Summer land, Calif.
www.americanradiohistory.com
HANDBOOK
Figure
Antenna Bandwidth
409
11
COMPARATIVE VERTICAL
RADIATION PATTERNS
Showing the vertical radiation
patterns of a horizontal single section flat -top beam (A), an
array of two stacked horizontal
-
half -wave elements
half of a "Lazy H "-(8), and
a horizontal dipole (C). In each
case the top of the antenna system is 0.75 wavelength above
ground, as shown to the left of
in -phase
the curves.
angle radiation at the expense of the useless
high -angle radiation with these simple arrays
as contrasted to the dipole is quite marked.
Figure 12 compares the patterns of a 3 element beam and a dipole radiator at a height of
0.75 wavelength. It will be noticed that although there is more energy in the lobe of the
beam as compared to the dipole, the axis of
the beam is at the same angle above the horizontal. Thus, although more radiated energy
is provided by the beam at low angles, the
average angle of radiation of the beam is no
lower than the average angle of radiation of
the dipole.
21 -5
21 -6
Propagation of
Radio Waves
The preceding sections have discussed the
manner in which an electromagnetic -wave or
radio -wave field may be set up by a radiating
system. However, for this field to be useful
for communication it must be propagated to
some distant point where it may be received,
or where it may be reflected so that it may be
received at some other point. Radio waves
may be propagated to a remote point by either
or both of two general methods. Propagation
Bandwidth
The bandwidth of an antenna or an antenna
array is a function primarily of the radiation
resistance and of the shape of the conductors
which make up the antenna system. For arrays
of essentially similar construction the bandwidth (or the deviation in frequency which the
system can handle without mismatch) is increased with increasing radiation resistance,
and the bandwidth is increased with the use
of conductors of larger diameter (smaller ratio
of length to diameter). This is to say that if
an array of any type is constructed of large
diameter tubing or spaced wires, its bandwidth
will be greater than that of a similar array
constructed of single wires.
The radiation resistance of antenna arrays
of the types mentioned in the previous paragraphs may be increased through the use of
wider spacing between elements. With increased
radiation resistance in such arrays the radiation efficiency increases since the ohmic
losses within the conductors become a smaller
percentage of the radiation resistance, and the
bandwidth is increased proportionately.
\
A- DIPOLE
B-3- ELEMENT
PARASITIC
0
1.5
2.0
2.5
3.0
GAIN IN FIELD STRENGTH
Figure
3.3
12
VERTICAL RADIATION PATTERNS
Showing vertical radiation patterns of a horizontal dipole (A) and a horizontal 3- element
parasitic array (8) at a height above ground
of 0.75 wavelength. Note that the axis of the
main radiation lobes are at the some angle
above the horizontal. Note also the suppression of high angle radiation by the parasitic
www.americanradiohistory.com
array.
408
RADIO
THE
Radiation, Propagation and Lines
Figure
9
VERTICAL RADIATION
PATTERNS
Showing the vertical radiation
patterns for half -wave antennas
(or colinear half -wave or extended half-wave antennas) at
different heights above average
ground and perfect ground. Note
that such antennas one -quarter
wave above ground concentrate
most radiation at the very high
angles which are useful for communication only on the lower frequency bands. Antennas one-half
wave above ground are not
shown, but the elevation pattern
shows one lobe on each side at
an
POWER OUTPUT
dipole could be increased by raising the antenna higher above the ground. This is true to
an extent in the case of the horizontal dipole;
the low -angle radiation does increase slowly
after a height of 0.6 wavelength is reached
but at the expense of greatly increased high angle radiation and the formation of a number
of nulls in the elevation pattern. No signal
can be transmitted or received at the elevation
angles where these nulls have been formed.
Tests have shown that a center height of 0.6
wavelength for a vertical dipole (0.35 wavelength to the bottom end) is about optimum for
this type of array.
Figure 9 shows the effect of placing a horizontal dipole at various heights above ground.
It is easily seen by reference to figure 9 (and
figure 10 which shows the radiation from a dipole at ja wave height) that a large percentage
of the total radiation from the dipole is being
radiated at relatively high angles which are
useless for communication on the 14 -Mc. and
28 -Mc. bands. Thus we see that in order to obtain a worthwhile increase in the ratio of low angle radiation to high -angle radiation it is
necessary to place the antenna high above
ground, and in addition it is necessary to use
angle of
30. above horizontal.
additional means for suppressing high -angle
radiation.
High -angle radiation can be
suppressed, and this radiation
can be added to that going out
at low angles, only through the
use of some sort of directive antenna system.
There are three general types of antenna arrays composed of dipole elements commonly
used which concentrate radiation at the lower
more effective angles for high -frequency communication. These types are: (1) The close spaced out -of -phase system as exemplified by
the "flat -top" beam or a8JK array. Such configurations are classified as end fire arrays.
(2) The wide - spaced in -phase arrays, as exemplified by the "Lazy H" antenna. These configurations are classified as broadside arrays.
(3) The close- spaced parasitic systems, as
exemplified by the three element rotary beam.
A comparison between the radiation from a
dipole, a "flat -top beam" and a pair of dipoles
stacked one above the other (half of a "lazy
H "), in each case with the top of the antenna
at a height of Sa wavelength is shown in figure
11. The improvement in the amplitude of lowSuppression of
High -angle
Radiation
Figure
10
VERTICAL RADIATION
PATTERNS
Showing vertical -plane radiation
patterns of a horizontal single section flat -top beam with one-
eighth wave spacing (solid
curves) and a horizontal halfantenna (dashed curves)
when both are 0.5 wavelength
(A) and 0.75 wavelength (B) a-
wav
.5
1.0
1.5
2.0
Q.0
3.0
.0
1.0
1.5
2.0
2.0
3.0
GAIN IN FIELD STRENGTH
www.americanradiohistory.com
bove ground.
Antenna
HANDBOOK
Directivity
407
radiated at other elevation angles is lost
and performs no useful function.
gy
The optimum angle of radiation
for propagation of signals between two points is dependent
upon a number of variables. Among these significant variables are: (1) height of the ionosphere layer which is providing the reflection,
(2) distance between the two stations, (3) number of hops for propagation between the two
stations. For communication on the 14 -Mc.
band it is often possible for different modes
of propagation to provide signals between two
points. This means, of course, that more than
one angle of radiation can be used. If no elevation directivity is being used under this condition of propagation, selective fading will
take place because of interference between the
waves arriving over the different paths.
On the 28 -Mc. band it is by far the most common condition that only one mode of propagation
will be possible between two points at any
one time. This explains, of course, the reason
why rapid fading in general and selective fading in particular are almost absent from signals heard on the 28 -Mc. band (except for fading caused by local effects).
Measurements have shown that the angles
useful for communication on the 14-Mc. band
are from 3° to about 30 °; angles above about
15° being useful only for local work. On the
28 -Mc. band measurements have shown that
the useful angles range from about 3° to 18 °;
angles above about 12° being useful only for
local (less than 3000 miles) work. These figures assume normal propagation by virtue of
the 1:2 layer.
Optimum Angle
of Radiation
.2
.4
.3
.2
.1
0
30
22
26 24
22 20 M N 14 12 IS
WAVE ANGLE IN DEGREES
2
0
Figure 8
DIRECTIONAL CHARACTERISTICS OF HORIZONTAL AND VERTICAL DOUBLETS ELEVATED 0.6 WAVELENGTH AND ABOVE TWO TYPES OF
VERTICALPLANE
GROUND
H, represents a horizontal doublet over typical farmland. H2 over salt water. VI is a
vertical pattern of radiation from o vertical
doublet over typical farmland, V2 over salt
water. A salt water ground is the closest
approach to an extensive ideally perfect
ground that will be met in actual practice.
great -circle path, or within 2 or 3 degrees of
that path under all normal propagation conditions. However, under turbulent ionosphere
conditions, or when unusual propagation conditions exist, the deviation from the great -circle
path for greatest signal intensity may be as
great as 90 °. Making the array rotatable overcomes these difficulties, but arrays having extremely high horizontal directivity become too
cumbersome to be rotated, except perhaps
when designed for operation on frequencies
above 50 Mc.
Vertical directivity is of the great est importance in obtaining satisfactory communication above 14
Mc. whether or not horizontal directivity is
used. This is true simply because only the
energy radiated between certain definite elevation angles is useful for communication. EnerVertical
Directivity
Angle of Radiation
of Typical Antennos
and Arrays
It now becomes of interest to determine the smount of radiation available at these useful lower angles of radiation from commonly used an-
tennas and antenna arrays. Figure 8 shows
relative output voltage plotted against elevation angle (wave angle) in degrees above the
horizontal, for horizontal and vertical doublets
elevated 0.6 wavelength above two types of
ground. It is obvious by inspection of the
curves that a horizontal dipole mounted at this
height above ground (20 feet on the 28 -Mc.
band) is radiating only a small amount of energy at angles useful for communication on the
28 -Mc. band. Most of the energy is being radiated uselessly upward. The vertical antenna
above a good reflecting surface appears much
better in this respect -and this fact has been
proven many times by actual installations.
It might immediately be thought that the amount of radiation from a horizontal or vertical
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406
is resistance of the wire, ground resistance
(in the case of a Marconi), corona discharge,
and insulator losses.
The approximate effective radiation efficiency (expressed as a decimal) is equal to:
Nr = Ra /(Ra+ RL) where R. is equal to the
radiation resistance and RL is equal to the
effective loss resistance of the antenna. The
loss resistance will be of the order of 0.25
ohm for large- diameter tubing conductors such
as are most commonly used in multi- element
parasitic arrays, and will be of the order of
0.5 to 2.0 ohms for arrays of normal construction using copper wire.
When the radiation resistance of an antenna
or array is very low, the current at a voltage
node will be quite high for a given power. Likewise, the voltage at a current node will be very
high. Even with a heavy conductor and excellent insulation, the losses due to the high voltage and current will be appreciable if the radiation resistance is sufficiently low.
Usually, it is not considered desirable to
use an antenna or array with a radiation resistance of less than approximately 5 ohms unless
there is sufficient directivity, compactness,
or other advantage to offset the losses resulting from the low radiation resistance.
The radiation resistance of a Marconi antenna, especially, should
be kept as high as possible. This
will reduce the antenna current for a given
power, thus minimizing loss resulting from the
series resistance offered by the earth connection. The radiation resistance can be kept high
by making the Marconi radiator somewhat longer
than a quarter wave, and shortening it by series
capacitance to an electrical quarter wave. This
reduces the current flowing in the earth connection. It also should be removed from ground
as much as possible (vertical being ideal).
Methods of minimizing the resistance of the
earth connection will be found in the discussion of the Marconi antenna.
Ground
Resistance
21 -4
THE
Radiation, Propagation and Lines
Antenna Directivity
All practical antennas radiate better in some
directions than others. This characteristic is
called directivity. The more directive an antenna is, the more it concentrates the radiation
in a certain direction, or directions. The more
the radiation is concentrated in a certain direction, the greater will be the field strength produced in that direction for a given amount of
total radiated power. Thus the use of a directional antenna or array produces the same result in the favored direction as an increase in
the power of the transmitter.
The increase in radiated power in a certain
RADIO
direction with respect to an antenna in free
space as a result of inherent directivity is
called the free space directivity power gain
or just space directivity gain of the antenna
(referred to a hypothetical isotropic radiator
which is assumed to radiate equally well in
all directions). Because the fictitious isotropic
radiator is a purely academic antenna, not physically realizable, it is common practice to use
as a reference antenna the simplest ungrounded resonant radiator, the half -wave Hertz, or
resonant doublet. As a half-wave doublet has
a space directivity gain of 2.15 db over an isotropic radiator, the use of a resonant dipole
as the comparison antenna reduces the gain
figure of an array by 2.15 db. However, it should
be understood that power gain can be expressed
with regard to any antenna, just so long as it
is specified.
As a matter of interest, the directivity of
an infinitesimal dipole provides a free space
directivity power gain of 1.5 (or 1.76 db) over
an isotropic radiator. This means that in the
direction of maximum radiation the infinitesimal dipole will produce the same field of
strength as an isotropic radiator which is radiating 1.5 times as much total power.
A half -wave resonant doublet, because of
its different current distribution and significant length, exhibits slightly more free space
power gain as a result of directivity than does
the infinitesimal dipole, for reasons which will
be explained in a later section. The space
directivity power gain of a half -wave resonant
doublet is 1.63 (or 2.15 db) referred to an isotropic radiator.
choosing and orienting an
antenna system, the radiation patterns of the various common types
of antennas should be given careful consideration. The directional characteristics are of
still greater importance when a directive antenna array is used.
Horizontal directivity is always desirable
on any frequency for point -to -point work. However, it is not always attainable with reasonable antenna dimensions on the lower frequencies. Further, when it is attainable, as
on the frequencies above perhaps 7 Mc., with
reasonable antenna dimensions, operating convenience is greatly furthered if the maximum
lobe of the horizontal directivity is controllable. It is for this reason that rotatable antenna arrays have come into such common
usage.
Considerable horizontal directivity can be
used to advantage when: (1) only point -topoint work is necessary, (2) several arrays are
available so that directivity may be changed
by selecting or reversing antennas, (3) a single
rotatable array is in use. Signals follow the
Horizontal
Directivity
When
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HANDBOOK
Antenna Impedance
HEIGHT IN WAVELENGTHS OF CENTER OF VERTICAL
HALF -WAVE ANTENNA ABOVE PERFECT GROUND
25
.3
.4
.5
.5
.7
.75
V
0
.1
w-_
571
.2
-MON OH7AL
.3
.4
.S
41
.7
.
.5
to
HEIGHT IN WAVELENGTHS OF HORIZONTAL HALF WAVE ANTENNA ABOVE PERFECT GROUND
Figure 7
EFFECT OF HEIGHT ON THE RADIATION
RESISTANCE OF A DIPOLE SUSPENDED
ABOVE PERFECT GROUND
the radiation resistance to approximately 100
ohms. When a horizontal half -wave antenna is
used, the radiation resistance (and, of course,
the amount of energy radiated for a given antenna current) depends on the height of the
antenna above ground, since the height determines the phase and amplitude of the wave
reflected from the ground back to the antenna.
Thus the resultant current in the antenna for
a given power is a function of antenna height.
linear radiator is series fed
the center, the resistive and
reactive components of the driving
point impedance are dependent upon both the length and diameter of the radiator
in wavelengths. The manner in which the resistive component varies with the physical dimensions of the radiator is illustrated in figure 5.
The manner in which the reactive component
varies is illustrated in figure 6.
Several interesting things will be noted with
respect to these curves. The reactive component disappears when the overall physical
length is slightly less than any number of half
waves long, the differential increasing with
conductor diameter. For overall lengths in the
vicinity of an odd number of half wavelengths,
the center feed point looks to the generator or
transmission line like a series -resonant lumped
circuit, while for overall lengths in the vicinity of an even number of half wavelengths, it
looks like a parallel- resonant or anti- resonant
lumped circuit. Both the feed point resistance
Center -fed
When a
Feed Point
Impedance
at
405
and the feed point reactance change more slowly with overall radiator length (or with frequency with a fixed length) as the conductor
diameter is increased, indicating that the effective "Q" is lowered as the diameter is in-
creased. However, in view of the fact that the
damping resistance is nearly all "radiation
resistance" rather than loss resistance, the
lower Q does not represent lower efficiency.
Therefore, the lower Q is desirable, because
it permits use of the radiator over a wider frequency range without resorting to means for
eliminating the reactive component. Thus, the
use of a large diameter conductor makes the
overall system less frequency sensitive. If the
diameter is made sufficiently large in terms of
wavelengths, the Q will be low enough to qualify the radiator as a "broad- band" antenna.
The curves of figure 7 indicate the theoretical center -point radiation resistance of a half wave antenna for various heights above perfect
ground. These values are of importance in
matching untuned radio -frequency feeders to
the antenna, in order to obtain a good impedance match and an absence of standing waves
on the feeders.
Ground Losses
Above
average ground, the
actual radiation resistance
of a dipole will vary from the exact value of
figure 7 since the latter assumes a hypothetical, perfect ground having no loss and perfect
reflection. Fortunately, the curves for the radiation resistance over most types of earth will
correspond rather closely with those of the
chart, except that the radiation resistance for
a horizontal dipole does not fall off as rapidly
as is indicated for heights below an eighth
wavelength. However, with the antenna so
close to the ground and the soil in a strong
field, much of the radiation resistance is actually represented by ground loss; this means
that a good portion of the antenna power is
being dissipated in the earth, which, unlike
the hypothetical perfect ground, has resistance.
In this case, an appreciable portion of the
radiation resistance actually is loss resistance. The type of soil also has an effect upon
the radiation pattern, especially in the vertical
plane, as will be seen later.
The radiation resistance of an antenna generally increases with length, although this increase varies up and down about a constantly
increasing average. The peaks and dips are
caused by the reactance of the antenna, when
its length does not allow it to resonate at the
operating frequency.
Antennas have a certain loss resistance as well as a radiation resistance. The loss resistance defines the power lost in the antenna due to ohmAntenna
Efficiency
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404
THE
Radiation, Propagation and Lines
10000
+9000
9000
+5000
8000
+4000
7000
+3000
6000
+2000
+ 1000
5000.
4000
DIAMETER=
rka000,
3000
2000
i
=r'
2000
l
DIAMETER1000
0
O.15Á
05A
1.05
1.5A
2.05
3000
4000
2.55
OVERALL LENGTH OF RADIATOR
Figure 5
FEED POINT RESISTANCE OF A CENTER
DRIVEN RADIATOR AS A FUNCTION OF
PHYSICAL LENGTH IN TERMS OF FREE
SPACE WAVELENGTH
When the antenna is resonant, and it always
should be for best results, the impedance at
the center is substantially resistive, and is
termed the radiation resistance. Radiation resistance is a fictitious term; it is that value
of resistance (referred to the current loop)
which would dissipate the same amount of
power as being radiated by the antenna, when
fed with the current flowing at the current loop.
The radiation resistance depends on the
antenna length and its proximity to nearby
objects which either absorb or re- radiate power, such as the ground, other wires, etc.
The Marconi
Antenna
RADIO
Before going too far with the
discussion of radiation resistance, an explanation of the Marconi (grounded quarter wave) antenna is in
order. The Marconi antenna is a special type
of Hertz antenna in which the earth acts as the
"other half" of the dipole. In other words, the
current flows into the earth instead of into a
similar quarter -wave section. Thus, the current
loop of a Marconi antenna is at the base rather
than in the center. In either case it is a quarter
wavelength from the end.
A half -wave dipole far from ground and other
reflecting objects has a radiation resistance
at the center of about 73 ohms. A Marconi an-
5000
0 155
055
OA
1.25
2.0
OVERALL LENGTH OF RADIATOR
I
2.55
Figure 6
REACTIVE COMPONENT OF THE FEED
A
CENTER
IMPEDANCE
OF
POINT
DRIVEN RADIATOR AS A FUNCTION OF
PHYSICAL LENGTH IN TERMS OF FREE
SPACE WAVELENGTH
tenna is simply one -half of a dipole. For that
reason, the radiation resistance is roughly
half the 73-ohm impedance of the dipole or
36.5 ohms. The radiation resistance of a Marconi antenna such as a mobile whip will be
lowered by the proximity of the automobile
body.
Because the power throughout the
antenna is the same, the impedance of a resonant antenna at any
point along its length merely expresses the
ratio between voltage and current at that point.
Thus, the lowest impedance occurs where the
current is highest, namely, at the center of a
dipole, or a quarter wave from the end of a
Antenna
Impedance
Marconi. The impedance rises uniformly toward
each end, where it is about 2000 ohms for a
dipole remote from ground, and about twice as
high for a vertical Marconi.
If a vertical half-wave antenna is set up so
that its lower end is at the ground level, the
effect of the ground reflection is to increase
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HANDBOOK
Radiation Resistance
A harmonic operated antenna is somewhat
longer than the corresponding integral number
of dipoles, and for this reason, the dipole
length formula cannot be used simply by multiplying by the corresponding harmonic. The
intermediate half wave sections do not have
end effects. Also, the current distribution is
disturbed by the fact that power can reach
some of the half wave sections only by flowing
through other sections, the latter then acting
not only as radiators, but also as transmission
lines. For the latter reason, the resonant length
will be dependent to an extent upon the method
of feed, as there will be less attenuation of
the current along the antenna if it is fed at or
near the center than if fed towards or at one
end. Thus, the antenna would have to be somewhat longer if fed near one end than if fed near
the center. The difference would be small,
however, unless the antenna were many wavelengths long.
The length of a center fed harmonically operated doublet may be found from the formula:
L
(K -.05) x 492
Freq. in Mc.
Under conditions of severe current attenuation, it is possible for some of the nodes, or
loops, actually to be slightly greater than a
physical half wavelength apart. Practice has
shown that the most practical method of resonating a harmonically operated antenna accurately is by cut and try, or by using a feed
system in which both the feed line and antenna
are resonated at the station end as an integral
system.
A dipole or half-wave antenna is said to
operate on its fundamental or first harmonic.
A full wave antenna, 1 wavelength long, operates on its second harmonic. An antenna with
five half- wavelengths on it would be operating
on its fifth harmonic. Observe that the fifth
harmonic antenna is 2tfs wavelengths long, not
wavelengths.
Antenna
Resonance
000
Di
o.sa rv=.
Figure
4
EFFECT OF SERIES INDUCTANCE AND
CAPACITANCE ON THE LENGTH OF A
HALF -WAVE RADIATOR
The top antenna has been electrically lengthened by placing o coil in series with the center. In other words, an antenna with a lumped
inductance in its center can be mode shorter
for a given frequency than a plain wire radiator. The bottom antenna has been capacitively shortened electrically. In other words, on
antenna with o capacitor in series with it
must be mode longer for o given frequency
since its effective electrical length os compared to plain wire is shorter.
i
where K = number of
waves on
antenna
L = length in feet
5
403
Most types of antennas operate
most efficiently when tuned or
resonated to the frequency of
operation. This consideration of course does
not apply to the rhombic antenna and to the
parasitic elements of arrays employing parasitically excited elements. However, in practically every other case it will be found that increased efficiency results when the entire antenna system is resonant, whether it be a simple dipole or an elaborate array. The radiation
efficiency of a resonant wire is many times
that of a wire which is not resonant.
If an antenna is slightly too long, it can be
resonated by series insertion of a variable
capacitor at a high current point. If it is slightly too short, it can be resonated by means of
a variable inductance. These two methods,
illustrated schematically in figure 4, are generally employed when part of the antenna is
brought into the operating room.
With an antenna array, or an antenna fed by
means of a transmission line, it is more common to cut the elements to exact resonant
length by "cut and try" procedure. Exact antenna resonance is more important when the
antenna system has low radiation resistance;
an antenna with low radiation resistance has
higher Q (tunes sharper) than an antenna with
high radiation resistance. The higher Q does
not indicate greater efficiency; it simply indicates a sharper resonance curve.
Radiation Resistance
21 -3
and Feed -Point Impedance
In many ways, a half-wave antenna is like a
tuned tank circuit. The main difference lies in
the fact that the elements of inductance, capacitance, and resistance are lumped in the tank
circuit, and are distributed throughout the
length of an antenna. The center of a half -wave
radiator is effectively at ground potential as
far as r-f voltage is concerned, although the
current is highest at that point.
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402
RADIO
THE
Radiation, Propagation and Lines
distance in meters between adjacent peaks
or adjacent troughs of a wave train.
As a radio wave travels 300,000,000 meters
a second (speed of light), a frequency of
cycle per second corresponds to a wavelength
1
of 300,000,000 meters. So, if the frequency is
multiplied by a million, the wavelength must
be divided by a million, in order to maintain
their correct ratio.
A frequency of 1,000,000 cycles per second
(1,000 kc.) equals a wavelength of 300 meters.
Multiplying frequency by 10 and dividing wavelength by 10, we find: a frequency of 10,000 kc.
equals a wavelength of 30 meters. Multiplying
and dividing by 10 again, we get: a frequency
of 100,000 kc. equals 3 meters wavelength.
Therefore, to change wavelength to frequency
(in kilocycles), simply divide 300,000 by the
wavelength in meters (À).
300,000
Fkc =
À
À
-
300,000
Fkc
Now that we have a simple conversion formula for converting wavelength to frequency
and vice versa, we can combine it with our
wavelength versus antenna length formula, and
we have the following:
Length of a half -wave radiator made from
wire (no. 14 to no. 10):
3.5 -11c. to 30 -Mc. bands
468
Length in feet
=
Freq. in Mc.
50 -Mc. band
Length in feet
Length in inches
-
40
.40
130
200
100
RATIO
OF
300 X00
400,000
400
shortening can be determined with the aid of
the chart of figure 3. In this chart the amount
of additional shortening over the values given
in the previous paragraph is plotted against
the ratio of the length to the diameter of the
half -wave radiator.
The length of a wave in free space is somewhat longer than the length of an antenna for
we same frequency. The actual free -space
half- wavelength is given by the following
expressions:
460
Freq. in Mc.
Half- wavelength =
5600
Freq. in Mc.
Half- wavelength
492
Freq. in Mc.
5905
w
5500
Mc.
half -wave radiator
is constructed from tubing
or rod whose diameter is
an appreciable fraction of the length of the
radiator, the resonant length of a half -wave
antenna will be shortened. The amount of
Length -to- Diameter
Ratio
When a
Mc.
in feet
in inches
wire in space can resonate at
more than one frequency. The lowest frequency at which it resonates
is called its fundamental frequency, and at
that frequency it is approximately a half wavelength long. A wire can have two, three, four,
five, or more standing waves on it, and thus
it resonates at approximately the integral harmonics of its fundamental frequency. However,
the higher harmonics are not exactly integral
multiples of the lowest resonant frequency as
a result of end effects.
Harmonic
Resonance
144 -Mc. band
Freq. in
.00
Figure 3
CHART SHOWING SHORTENING OF A
ELEMENT IN TERMS OF
RESONANT
RATIO OF LENGTH TO DIAMETER
The use of this chart is based on the basic
formula where radiator length in feet is
equal to 468 /frequency in Mc. This formula
applies to frequencies below perhaps 30 Mc.
when the radiator is made from wire. On
higher frequencies, or on 14 and 28 Mc. when
the radiator is made of large- diameter tubing,
the radiator is shortened from the value obtained with the above formula by on amount
determined by the ratio of length to diameter
of the radiator. The amount of this shortening is obtainable from the chart shown above.
Freq. in
Length in inches =
:000
LENGTH TO DIAMETER
A
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HANDBOOK
Antenna Characteristics
401
Figure 2
ANTENNA POLARIZATION
The polarization (electric field) of
the radiation from a resonant dipole
such as shown at (A) above is parallel to the length of the radiator. In
the case of o resonant slot cut in a
sheet of metal and used as a radiator, the polarization (of the electric field) is perpendicular to the
length of the slot. In both cases,
however, the polarization of the
radiated field is parallel to the potential gradient of the radiator; in
the case of the dipole the electric
lines of force are from end to end,
while in the case of the slot the
field is across the sides of the
slot. The metallic sheet containing
the slot may be formed into a cylinder to make up the radiator shown
at (C). With this type of radiator
the radiated field will be horizontally polarized even though the
radiator is mounted vertically.
is a graph showing the relative radiated field
intensity against azimuth angle for horizontal
directivity and field intensity against elevation
angle for vertical directivity.
The bandwidth of an antenna is a measure
of its ability to operate within specified limits
over a range of frequencies. Bandwidth can
be expressed either "operating frequency plus or -minus a specified per cent of operating frequency" or "operating frequency plus -or -minus
a specified number of megacycles" for a certain standing- wave -ratio limit on the transmission line feeding the antenna system.
The effective power gain or directive gain
of an antenna is the ratio between the power
required in the specified antenna and the power
required in a reference antenna (usually a halfwave dipole) to attain the same field strength
in the favored direction of the antenna under
measurement. Directive gain may be expressed
either as an actual power ratio, or as is more
common, the power ratio may be expressed
in decibels.
Physical Length
of a Half -Wave
cross section of the
conductor which makes up
Antenna
the antenna is kept very
small with respect to the
antenna length, an electrical half wave is a
fixed percentage shorter than a physical halfwavelength. This percentage is approximately
5 per cent. Therefore, most linear half-wave antennas are close to 95 per cent of a half wavelength long physically. Thus, a half-wave antenna resonant at exactly 80 meters would be
one -half of 0.95 times 80 meters in length. Another way of saying the same thing is that a
If the
ELECTRIC
FIELD
(POLARIZATION)
VERTICAL
O
ELECTRIC FIELD
(POLARIZATION)
HORIZONTAL
.or
FEEDERS CONNECT
TO POINTS Aas
NSIDC CYLINDER
wire resonates at a wavelength of about 2.1
times its length in meters. If the diameter of
the conductor begins to be an appreciable fraction of a wavelength, as when tubing is used
as a v -h -f radiator, the factor becomes slightly
less than 0.95. For the use of wire and not
tubing on frequencies below 30 Mc., however,
the figure of 0.95 may be taken as accurate.
This assumes a radiator removed from surrounding objects, and with no bends.
Simple conversion into feet can be obtained
by using the factor 1.56. To find the physical
length of a half -wave 80 -meter antenna, we
multiply 80 times 1.56, and get 124.8 feet for
the length of the radiator.
It is more common to use frequency than
wavelength when indicating a specific spot in
the radio spectrum. For this reason, the relationship between wavelength and frequency
must be kept in mind. As the velocity of radio
waves through space is constant at the speed
of light, it will be seen that the more waves
that pass a point per second(higher frequency),
the closer together the peaks of those waves
must be (shorter wavelength). Therefore, the
higher the frequency, the lower will be the
wavelength.
A radio wave in space can be compared to
a wave in water. The wave, in either case, has
peaks and troughs. One peak and one trough
constitute a full wave, or one wavelength.
Frequency describes the number of wave
cycles or peaks passing a point per second.
Wavelength describes the distance the wave
travels through space during one cycle or
oscillation of the antenna current; it is the
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THE
Radiation, Propagation and Lines
400
`
VOLTAGE
. ,---t--_
me/m..1.5,-k\
C[NTCII
'
` `{
.
1...-14ALW-WAVE ANTENNA
f
i
.
SHOWING NOW STANDING WAVES
CRUST ON A HORIZONTAL ANTENNA.
C.
VOLTAGE
CURRENT IS MAXIMUM AT CENTRA.
VOLTAGE IS MAXIMUM AT
Figure
STANDING WAVES ON A RESONANT
ANTENNA
1
transmission lines, both from single -wire lines
and from lines comprised of more than one
wire. In addition, radiation can be made to
take place in a very efficient manner from electromagnetic horns, from plastic lenses or from
electromagnetic lenses made up of spaced conducting planes, from slots cut in a piece of
metal, from dielectric wires, or from the open
end of a wave guide.
Directivity of
Radiation
The radiation from any phys-
ically practicable radiating
system is directive to a certain
degree. The degree of directivity can be enhanced or altered when desirable through the
combination of radiating elements in a prescribed manner, through the use of reflecting
planes or curved surfaces, or through the use
of such systems as mentioned in the preceding
paragraph. The construction of directive antenna arrays is covered in detail in the chapters which follow.
Like light waves, radio waves
can have a definite polarization.
In fact, while light waves ordinarily have to
be reflected or passed through a polarizing
medium before they have a definite polarization, a radio wave leaving a simple radiator
will have a definite polarization, the polarization being indicated by the orientation of
the electric -field component of the wave. This,
in turn, is determined by the orientation of the
radiator itself, as the magnetic -field component
is always at right angles to a linear radiator,
and the electric -field component is always in
the same plane as the radiator. Thus we see
that an antenna that is vertical with respect
to the earth will transmit a vertically polarized wave, as the electrostatic lines of force
will be vertical. Likewise, a simple horizontal
antenna will radiate horizontally polarized
waves.
Polarization
RADIO
Because the orientation of a simple linear
radiator is the same as the polarization of the
waves emitted by it, the radiator itself is referred to as being either vertically or horizontally polarized. Thus, we say that a horizontal
antenna is horizontally polarized.
Figure 2A illustrates the fact that the polarization of the electric field of the radiation
from a vertical dipole is vertical. Figure 2B,
on the other hand, shows that the polarization
of electric -field radiation from a vertical slot
radiator is horizontal. This fact has been utilized in certain commercial FM antennas where
it is desired to have horizontally polarized
radiation but where it is more convenient to
use an array of vertically stacked slot arrays.
If the metallic sheet is bent into a cylinder
with the slot on one side, substantially omnidirectional horizontal coverage is obtained
with horizontally -polarized radiation when the
cylinder with the slot in one side is oriented
vertically. An arrangement of this type is shown
in figure 2C. Several such cylinders may be
stacked vertically to reduce high -angle radiation and to concentrate the radiated energy
at the useful low radiation angles.
In any event the polarization of radiation
from a radiating system is parallel to the electric field as it is set up inside or in the vicinity of the radiating system.
21 -2
General Character-
istics of Antennas
antennas have certain general characterIt is the result of
differences in these general characteristics
which makes one type of antenna system most
suitable for one type of application and another type best for a different application. Six
All
istics to be enumerated.
of the more important characteristics are: (1)
polarization, (2) radiation resistance, (3) horizontal directivity, (4) vertical directivity,
(5) bandwidth, and (6) effective power gain.
The polarization of an antenna or radiating
system is the direction of the electric field
and has been defined in Section 21 -1.
The radiation resistance of an antenna system is normally referred to the feed point in
an antenna fed at a current loop, or it is referred to a current loop in an antenna system
fed at another point. The radiation resistance
is that value of resistance which, if inserted
in series with the antenna at a current loop,
would dissipate the same energy as is actually
radiated by the antenna if the antenna current
at the feed point were to remain the same.
The horizontal and vertical directivity can
best be expressed as a directive pattern which
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CHAPTER TWENTY -ONE
Radiation, Propagation
and Transmission Lines
Radio waves are electromagnetic waves
similar in nature but much lower in frequency
than light waves or heat waves. Such waves
represent electric energy traveling through
space. Radio waves travel in free space with
the velocity of light and can be reflected and
refracted much the same as light waves.
21 -1
Radiation from
possible change in the electrical constants
of a line is that which occurs at the open end
of a wire. Therefore, a dipole has a great mismatch at each end, producing a high degree of
reflection. We say that the ends of a dipole
are terminated in an infinite impedance.
A returning wave which has been reflected
meets the next incident wave, and the voltage
and current at any point along the antenna are
the vector sum of the two waves. At the ends
of the dipole, the voltages add, while the currents of the two waves cancel, thus producing
high voltage and low current at the ends of the
dipole or half wave section of wire. In the
same manner, it is found that the currents add
while the voltages cancel at the center of the
dipole. Thus, at the center there is high current but low voltage.
Inspection of figure 1 will show that the
current in a dipole decreases sinusoidally
towards either end, while the voltage similarly
increases. The voltages at the two ends of the
antenna are 180° out of phase, which means
that the polarities are opposite, one being plus
while the other is minus at any instant. A
curve representing either the voltage or current on a dipole represents a standing wave
on the wire.
an
Antenna
Alternating current passing through a conductor creates an alternating electromagnetic
field around that conductor. Energy is alternately stored in the field, and then returned
to the conductor. As the frequency is raised,
more and more of the energy does not return
to the conductor, but instead is radiated off
into space in the form of electromagnetic
waves, called radio waves. Radiation from a
wire, or wires, is materially increased whenever there is a sudden change in the electrical
constants of the line. These sudden changes
produce reflection, which places standing
waves on the line.
When a wire in space is fed radio frequency
energy having a wavelength of approximately
2.1 times the length of the wire in meters, the
wire resonates as a half-wave dipole antenna
at that wavelength or frequency. The greatest
Radiation from
Sources other
than Antennas
399
www.americanradiohistory.com
Radiation can and does take
place from sources other than
antennas. Undesired radiation
can take place from open -wire
3
98
OSCILLATOR
BUFFER
i
BUFFER Sta
I
+300 V.
OUTPUT
CONTROL
°
RI
100E
2w
-120V
Figure
19
DIFFERENTIAL KEYING SYSTEM WITH
OSCILLATOR SWITCHING DIODE
Vi
V2
OSCILLATOR
V3
DRIVER
BUFFER
300
V.
n
°
V4
12AU7
100 It
REVER TUBE
6
22
REV
005
100 E
R2
R3
100E
4TE
VFO"MOLD
-50V
005
100E
330E
C11`OS
Figure
20
DIFFERENTIAL KEYER EMPLOYED
IN
"JOHNSON" TRANSMITTERS
conducting--and then continue operating
until atter V2 and V3 have stopped conducting. Potentiometer R1 adjusts the "hold"
time for VFO operation after the key is opened.
This
may be adjusted to cut off the VFO
between marks of keyed characters, thus
allowing rapid break -in operation.
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Differential
HANDBOOK
Keying
6AL5
u
BLOCKING DIODES
o
397
TO CATHODE
CIRCUIT OF
KEYED STAGE
Ó
O
O
f r
u
2
o
VI
-250
170
Pt
V.
VACUUM
TUBE
KEYER
U
ñ
(FIG. Po)
1
ó
-Y-CUr-OFF VALUE
AMPLIFIER
I
\
CUT -OFF
VALUE
OSC.
Figure
BLOCKING DIODES EMPLOYED
TO VARY TIME CONSTANT OF
"MAKE" AND "BREAK" CHARACTERISTICS OF VACUUM TUBE
KEYER
-- DURING DEPRESSED
KEY iS
THIS TIME
6-TRANSMITTER IS YON THE
AIR- DURING THIS TIME
Figure
18
17
TIME SEQUENCE OF A
DIFFERENTIAL KEYER
on a moment before the rest of the stages are
energized, and remains on a moment longer
than the other stages. The "chirp" or frequency shift associated with abrupt switching of
the oscillator is thus removed from the emitted
signal. In addition, the differential keyer can
apply waveshaping to the amplifier section
of the transmitter, eliminating the "click"
caused by rapid keying of the latter stages.
The ideal keying system would perform as
illustrated in figure 17. When the key is
closed, the oscillator reaches maximum output almost instantaneously. The following
stages reach maximum output in a fashion
determined by the waveshaping circuits of
the keyer. When the key is released, the output of the amplifier stages starts to decay
in a predetermined manner, followed shortly
thereafter by cessation of the oscillator. The
overall result of these actions is to provide
relatively soft "make" and "break" to the
keyed signal, meanwhile preventing oscillator frequency shift during the keying sequence.
The rates of charge and decay in a typical
R -C keying circuit may be varied independently of each other by the blocking diode system
of figure 18. Each diode permits the charging
current of the timing capacitor to flow through
only one of the two variable potentiometers,
thus permitting independent adjustment of
the "make" and "break" characteristics of
the keying system.
A practical differential keying system de-
veloped by WIICP (Feb., 1956 QST) is shown
in figure 19. A 6AL5 switch tube turns the
oscillator on before the keying action starts,
and holds it on until after the keying sequence is completed. Time constant of the
keying cycle is determined by values of C
and R. When the key is open, a cut -off bias
of about -110 volts is applied to the screen
grid circuits of the keyed stages. When the
key is closed, the screen grid voltage rises
to the normal value at a rate determined by
the time constant R -C. Upon opening the key
again, the screen voltage returns to cut -off
value at the predetermined rate.
The potentiometer R1 serves as an output
control, varying the minimum internal resistance of the 12BH7 keyer tube, and is a
useful device to limit power input during tune up periods. Excitation to the final amplifier
stage may be controlled by the screen potentiometer R3 in the second buffer stage.
An external bias source of approximately -120
volts at 10 milliamperes is required for operation of the keyer, in addition to the 300-volt
screen supply.
Blocking voltage may be removed from the
oscillator for "zeroing" purposes by closing
switch Si, rendering the diode switch in-
operative.
keying system is shown
figure 20, and is widely used in many
Johnson transmitters. Grid block keying is
used on tubes V2 and V3. A waveshaping
filter consisting of R2, R3, and C1 is used
in the keying control circuit of V2 and V3.
To avoid chirp when the oscillator (V1) is
keyed, the keyer tube V4 allows the oscillator
to start quickly
before V2 and V3 start
A second popular
in
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--
396
Transmitter Keying and Control
THE
RADIO
LOW POWER SUFFER
6AG7
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KEYER UNIT
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VOLTS
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TIME
OUTPUT TO SCREEN CIS 807
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Figure 16
SCREEN GRID KEYER
guished, removing the screen voltage from the
tetrode r -f tube. At the same time, rectified
grid bias is applied to the screen of the tetrode
through the I megohm resistor between screen
and key. This voltage effectively cuts off the
screen of the tetrode until the key is closed
again. The RC circuit in the grid of the 6L6
tube determines the keying characteristic of
the tetrode tube.
A more elaborate screen grid keyer is shown
in figures 15 and 16. This keyer is designed
to block -grid key the oscillator or a low powered buffer stage, and to screen key a medium
powered tetrode tube such as an 807, 2E26 or
6146. The unit described includes a simple
dual voltage power supply for the positive
screen voltage of the tetrode, and a negative
supply for the keyer stages. A 6K6 is used as
the screen keyer, and a 12AU7 is used as a
cathode follower and grid block keyer. As in
UNIT
the W1DX keyer, this keyer turns on the exciter a moment before the tetrode stage is
turned on. The tetrode stage goes off an instant before the exciter does. Thus any keying chirp of the oscillator is effectively removed from the keyed signal.
By listening in the receiver one can hear
the exciter stop operating a fraction of a second after the tetrode stage goes off. In fact,
during rapid keying, the exciter may be heard
as a steady signal in the receiver, as it has
appreciable time lag in the keying circuit. The
clipping effect of following stages has a definite hardening effect on this, however.
20 -8
Differential Keying Circuits
Excellent waveshaping may be obtained by
differential keying system whereby the
master oscillator of the transmitter is turned
a
www.americanradiohistory.com
Screen
HANDBOOK
807.
Keying
EXC.
20.7
14
SINGLE -STAGE SCREEN GRID KEYER
FOR TETRODE TUBES
tetrode is keyed by this method, there is the
possibility of a considerable backwave caused
by r -f leakage through the grid -plate capacity
of the tube.
Certain hi -µ triode tubes, such as the 811 -A
and the 805, automatically block themselves
when the grid return circuit is opened. It is
merely necessary to insert a key and associated key click filter in the grid return lead of
these tubes. No blocking bias supply is needed. This circuit is shown in figure 12.
A more elaborate blocked -grid keying system has been developed by W1DX, and was
shown in the February, 1954 issue of QST
magazine. This highly recommended circuit
is shown in figure 13. Two stages are keyed,
Screen Grid Keying
The screen circuit of a tetrode tube may be
keyed for c -w operation. Unfortunately, when
the screen grid of a tetrode tube is brought to
zero potential, the tube still delivers considerable output. Thus it is necessary to place
a negative blocking voltage on the screen grid
to reduce the backwave through the tube. A
suitable keyer circuit that will achieve this
was developed by W6DTY, and was described
in the February, 1953 issue of CQ magazine.
This circuit is shown in figure 14. A 6L6 is
used as a combined clamper tube and keying
tube. When the key is closed, the 6L6 tube
has blocking bias applied to its control grid.
This bias is obtained from the rectified grid
bias of the keyed tube. Screen voltage is applied to the keyed stage through a screen dropping resistor and a VR -105 regulator tube.
then the key is open, the 6L6 is no longer
cut -off, and conducts heavily. The voltage
drop across the dropping resistor caused by
the heavy plate current of the 6L6 lowers the
voltage on the VR -105 tube until it is extin-
I^N
Figure
15
TOP VIEW OF SCREEN
GRID KEYER SHOWN IN
FIGURE
395
preventing any backwave emission. The first
keyed stage may be the oscillator, or a low
powered buffer. The last keyed stage may be
the driver stage to the power amplifier, or the
amplifier itself. Since the circuit is so proportioned that the lower powered stage comes
on /first and goes off last, any keying chirp in
the oscillator is not emitted on the air. Keying
lag is applied to the high powered keyed stage
only.
ETC
Figure
Grid
16
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Transmitter Keying
394
THE
and Control
RADIO
HI -MU TRIODE
(61 -A ETC.)
I
H.V
-BLOCKING
B AS
Figure 11
SIMPLE BLOCKED -GRID KEYING
SYSTEM
The blocking bias must be sufficient to cutoff plate current to the amplifier stage in the
presence of the excitation voltage. R¡ is normal bias resistor for the tube. R2 and C1
should be adjusted for correct keying waveform.
recommended for general use, as considerable
voltage will be developed across the key when
it is open.
An electronic switch can take the place of
the hand key. This will remove the danger of
shock. At the same time, the opening and closing characteristics of the electronic switch
may easily be altered to suit the particular
need at hand. Such an electronic switch is
called a vacuum tube keyer. Low internal resistance triode tubes such as the 45, 6A3, or
6AS7 are used in the keyer. These tubes act
as a very high resistance when sufficient
807, 6146,
ISO
LOW POWER BUFFER
(5457 ETC.)
ETC.
LUF
RFC
2
SUN
33K
2W
1001t
IW
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Figure 12
SELF -BLOCKING KEYING SYSTEM FOR
HIGH -MU TRIODE
R, and C1 adjusted for correct keying waveform. R, is bias resistor of tube.
blocking bias is applied to them, and as a
very low resistance when the bias is removed.
The desired amount of lag or cushioning effect
can be obtained by employing suitable resistance and capacitance values in the grid of the
keyer tube(s). Because very little spark is
produced at the key, due to the small amount
of power in the key circuit, sparking clicks
are easily suppressed.
One type 45 tube should be used for every
50 ma. of plate current. Type 6B4G or 2A3
tubes may also be used; allow one 6B4G tube
for every 80 ma. of plate current.
Because of the series resistance of the keyer
tubes, the plate voltage at the keyed tube will
be from 30 to 60 volts less than the power
supply voltage. This voltage appears as cathode bias on the keyed tube, assuming the bias
return is made to ground, and should be taken
into consideration when providing bias.
Some typical cathode circuit vacuum tube
keying units are shown in figure 10.
V.
VR-150
REY
-+00
6.3V.
TO
V.
6J5
Figure 13
BLOCKED-GRID KEYER
A separate filament transformer must be used
for the 6J5, as its filament is at a potential
of -400 volts.
TWO -STAGE
20-6
Grid Circuit Keying
Grid circuit, or blocked grid keying is another effective method of keying a c -w transmitter. A basic blocked grid keying circuit is
shown in figure 11. The time constant of the
keying is determined by the RC circuit, which
also forms part of the bias circuit of the tube.
When the key is closed, operating bias is developed by the flow of grid current through 121.
When the key is open, sufficient fixed bias is
applied to the tube to block it, preventing the
stage from functioning. If an un- neutralized
HANDBOOK
Cathode
TO
SOMA SELENIUM RECTIFIER
7.
I
6Y6
IM (BREAK)
I
M(MAKE)
Keying
393
RF TUBE
CATHODE OF
KEYED STAGE
I
W
STANCOR PAB421
45/2A3
2K ,2W
350.0 -350
70
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Figure
10
VACUUM TUBE KEYERS FOR CENTER -TAP
KEYING CIRCUITS
The type A keyer is suitable for keying stages
running up to 1250 volts on the plate. Two 2A3
or 6A3 tubes can safely key 160 milliamperes
of cathode current. The simple 6Y6 keyer in figure B Is for keying stages running up to 650 volts
on the plate. A single 6Y6 can key 80 milliamperes. Two in parallel may be used for plate
currents under 160 ma. If softer keying is desired, the 500 -µofd. mica condenser should be increased
to .001 pfd.
amplifier. If a low -level stage, which is followed by a series of class C amplifiers, is
keyed, serious transients will be generated
in the output of the transmitter even though
the keyed stage is being turned on and off very
smoothly. This condition arises as a result of
pulse sharpening, which has been discussed
previously.
Third, the output from the stage should be
completely cut off when the key is up, and the
time constant of the rise and decay of the keying wave should be easily controllable.
Fourth, it should be possible to make the
rise period and the decay period of the keying
wave approximately equal. This type of keying
envelope is the only one tolerable for commercial work, and is equally desirable for obtaining clean cut and easily readable signals in
amateur work.
Fifth, it is desirable that the keying circuit
be usable without a keying relay, even when
a high -power stage is being keyed.
Last, for the sake of simplicity and safety,
it should be possible to ground the frame of
the key, and yet the circuit should be such
that placing the fingers across the key will
not result in an electrical shock. In other words,
the keying circuit should be inherently safe.
All these requirements have been met in the
keying circuits to be described.
20-5
Cathode Keying
The lead from the cathode or center -tap connection of the filament of an r -f amplifier can
be opened and closed for a keying circuit. Such
a keying system opens the plate voltage circuit and at the same time opens the grid bias
return lead. For this reason, the grid circuit
is blocked at the same time the plate circuit
is opened. This helps to reduce the backwave
that might otherwise leak through the keyed
stage.
The simplest cathode keying circuit is illustrated in figure 9, where a key -click filter
is employed, and a hand key is used to break
the circuit. This simple keying circuit is not
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392
THE
Transmitter Keying and Control
RADIO
a wide frequency band as sidebands and are
heard as clicks.
The cure for transient key clicks is relatively simple, although one would not believe
it, judging from the hordes of clicky, "snappy"
signals heard on the air.
To be capable of transmitting code characters and at the same time not splitting the
eardrums of neighboring amateurs, the c -w
transmitter MUST meet two important specifications.
1- It must have no parasitic oscillations
either in the stage being keyed or in any
succeeding stage.
2- It must have some device in the keying
circuit capable of shaping the leading
and trailing edge of the waveform.
Both these specifications must be met be/ore the transmitter is capable of c-w operation. Merely turning a transmitter on and off
by the haphazard insertion of a telegraph key
in some power lead is an invitation to trouble.
The two general methods of keying a transmitter are those which control the excitation
to the keyed amplifier, and those which control the plate or screen voltage applied to the
keyed amplifier.
Key -Click
Elimination
Key -click elimination is accomplished by preventing a too -rapid
make- and -break of power to the
antenna circuit, rounding off the keying characters so as to limit the sidebands to a value
which does not cause interference to adjacent
channels. Too much lag will prevent fast keying, but fortunately key clicks can be practically eliminated without limiting the speed
of manual (hand) keying. Some circuits which
eliminate key clicks introduce too much time lag and thereby add tails to the dots. These
tails may cause the signals to be difficult to
copy at high speeds.
Considerable thought should be
given as to which stage in a
transmitter is the proper one to
key. If the transmitter is keyed in a stage close
to the oscillator, the change in r-f loading of
the oscillator will cause the oscillator to shift
frequency with keying. This will cause the
signal to have a distinct chirp. The chirp will
be multiplied as many times as the frequency
of the oscillator is multiplied. A chirpy oscillator that would be passable on 80 meters
would be unusable on 28 Mc. c.w.
Keying the oscillator itself is an excellent
way to run into keying difficulties. If no key
click filter is used in the keying circuit, the
transmitter will have bad key clicks. If a key
click filter is used, the slow rise and decay
of oscillator voltage induced by the filter action will cause a keying chirp. This action is
Location of
Keyed Stage
O
IS
Figure
CENTER -TAP KEYING WITH CLICK
9
FILTER
The constants shown above are suggested as
starting values; considerable variation in
these values can be expected for optimum
keying of amplifiers of different operating
conditions. It is suggested that a keying relay be substituted for the key in the circuit
above wherever practicable.
true of all oscillators, whether electron coupled
or crystal controlled.
The more amplifier or doubler stages that
follow the keyed stage, the more difficult it is
to hold control of the shape of the keyed waveform. A heavily excited doubler stage or class
C stage acts as a peak clipper, tending to
square up a rounded keying impulse, and the
cumulative effect of several such stages cascaded is sufficient to square up the keyed
waveform to the point where bad clicks are
reimposed on a clean signal.
A good rule of thumb is to never key back
farther than one stage removed from the final
amplifier stage, and never key closer than one
stage removed from the frequency controlling
oscillator of the transmitter. Thus there will
always be one isolating stage between the
keyed stage and the oscillator, and one isolating stage between the keyed stage and the
antenna. At this point the waveform of the
keyed signal may be most easily controlled.
first place it may be established that the majority of
new design transmitters, and
many of those of older design as well, use a
medium power beam tetrode tube either as the
output stage or as the exciter for the output
stage of a high power transmitter. Thus the
transmitter usually will end up with a tube
Keyer Circuit
Requirements
In the
such as type 2E26, 807, 6146, 813, 4 -65A,
4E27/257B, 4 -125A or similar, or one of these
tubes will be used as the stage just ahead of
the output stage.
Second, it may be established that it is undesirable to key further down in the transmitter
chain than the stage just ahead of the final
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HANDBOOK
Transmitter Keying
For 100 per cent protection, just obey the
following rule: never work on the transmitter
or reach inside any protective cover except
when the green pilots are glowing. To avoid
confusion, no other green pilots should be used
on the transmitter; if you want an indicator
jewel to show when the filaments are lighted,
use amber instead of green.
Filter capacitors of good quality hold their charge for some
time, and when the voltage is more than 1000
volts it is just about as dangerous to get across
an undischarged 4 -pfd. filter capacitor as it is
to get across a high -voltage supply that is
turned on. Most power supplies incorporate
bleeders to improve regulation, but as these
are generally wire -wound resistors, and as
wire -wound resistors occasionally open up
without apparent cause, it is desirable to incorporate an auxiliary safety bleeder across
each heavy -duty bleeder. Carbon resistors will
not stand much dissipation and sometimes
change in value slightly with age. However,
the chance of their opening up when run well
within their dissipation rating is very small.
To make sure that all capacitors are bled, it
is best to short each one with an insulated
screwdriver. However, this is sometimes awkward and always inconvenient. One can be virtually sure by connecting auxiliary carbon
bleeders across all wire -wound bleeders used
on supplies of 1000 volts or more. For every
500 volts, connect in series a 500,000 -ohm
1 -watt carbon resistor. The drain will be negligible (1 ma.) and each resistor will have to
dissipate only 0.5 watt. Under these conditions the resistors will last indefinitely with
little chance of opening up. For a 1500-volt
supply, connect three 500,000 -ohm resistors in
series. If the voltage exceeds an integral number of 500 volt divisions, assume it is the next
higher integral value; for instance, assume
1800 volts as 2000 volts and use four resistors.
Do not attempt to use fewer resistors by
using a higher value for the resistors; not over
500 volts should appear across any single
1 -watt resistor.
In the event that the regular bleeder opens
up, it will take several seconds for the auxiliary bleeder to drain the capacitors down to a
safe voltage, because of the very high resistance. Therefore, i t is best to allow 10 or 15
seconds after turning off the plate supply before attempting to work on the transmitter.
If a 0 -1 d-c milliammeter is at hand, it may
be connected in series with the auxiliary
bleeder to act as a high voltage voltmeter.
Safety Bleeders
"Hot" Adjustments
Some amateurs contend
that it is almost impossible
391
to make certain adjustments, such as coupling
and neutralizing, unless the transmitter is running. The best.thing to do is to make all neutralizing and coupling devices adjustable from
the front panel by means of flexible control
shafts which are broken with insulated couplings to permit grounding of the panel bearing.
If your particular transmitter layout is such
that this is impracticable and you refuse to
throw the main switch to make an adjustment
-throw the main switch -take a reading -throw
the main switch-make an adjustment -and so
on, then protect yourself by making use of long
adjusting rods made from 1/-inch dowel sticks
which have been wiped with oil when perfectly
free from moisture.
If you are addicted to the use of pickup loop
and flashlight bulb as a resonance and neutralizing indicator, then fasten it to the end of a
long dowel stick and use it in that manner.
Protective Interlocks
increasing tendency toward construcWith the
tion of transmitters in enclosed steel cabinets
a transmitter becomes a particularly lethal device unless adequate safety provisions have
been incorporated. Even with a combined safety
signal and switch as shown in figure 8 it is
still conceivable that some person unfamiliar
with the transmitter could come in contact with
high voltage. It is therefore recommended that
the transmitter, wherever possible, be built
into a complete metal housing or cabinet and
that all doors or access covers be provided
with protective interlocks (all interlocks must
be connected in series) to remove the high
voltage whenever these doors or covers are
opened. The term "high voltage" should mean
any voltage above approximately 150 volts,
although it is still possible to obtain a serious
burn from a 150 -volt circuit under certain circumstances. The 150 -volt limit usually will
mean that grid -bias packs as well as high voltage packs should have their primary circuits opened when any interlock is opened.
20 -4
Transmitter Keying
The carrier from a c -w telegraph transmitter
must be broken into dots and dashes for the
transmission of code characters. The carrier
signal is of constant amplitude while the key
is closed, and is entirely removed when the
key is open. When code characters are being
transmitted, the carrier may be considered as
being modulated by the keying. If the change
from the no- output condition to full -output, or
vice versa, occurs coo rapidly, the rectangular
pulses which form the keying characters contain high- frequency components which take up
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390
THE
Transmitter Keying and Control
sary chances. However, no one is infallible,
and chances of an accident are greatly lessened if certain factors are taken into consideration in the design of a transmitter, in order
to protect the operator in the event of a lapse
of caution. If there are too many things one
must "watch out for" or keep in mind there is
a good chance that sooner or later there will
be a mishap; and it only takes one. When designing or constructing a transmitter, the following safety considerations should be given
attention.
For the utmost in protection, everything of metal on the front panel of
a transmitter capable of being touched by the
operator should be at ground potential. This
includes dial set screws, meter zero adjuster
screws, meter cases if of metal, meter jacks,
everything of metal protruding through the front
panel or capable of being touched or nearly
touched by the operator. This applies whether
or not the panel itself is of metal. Do not rely
upon the insulation of meter cases or tuning
knobs for protection.
The B negative or chassis of all plate power
supplies should be connected together, and to
an external ground such as a waterpipe.
Grounds
It is not necessary to resort
to rack and panel construction in order to provide complete enclosure of all components and wiring
of the transmitter. Even with metal- chassis
construction it is possible to arrange things so
as to incorporate a protective shielding housing which will not interfere with ventilation
yet will prevent contact with all wires and
components carrying high voltage d.c. or a.c.,
in addition to offering shielding action.
If everything on the front panel is at ground
potential (with respect to external ground) and
all units are effectively housed with protective
covers, then there is no danger except when
the operator must reach into the interior part
of the transmitter, as when changing coils,
neutralizing, adjusting coupling, or shooting
trouble. The latter procedure can be made safe
by making it possible for the operator to be
absolutely certain that all voltages have been
turned off and that they cannot be turned on
either by short circuit or accident. This can be
done by incorporation of the following system
of main primary switch and safety signal lights.
Exposed Wires
and Components
The common method of
using red pilot lights to
show when a circuit is on
is useless except from an ornamental standpoint. When the red pilot is not lit it usually
means that the circuit is turned off, but it can
Combined Safety
Signal and Switch
RADIO
3V TO GREEN PILOT LIGHTS ON
FRONT PANEL AND ON EACH CHASSIS
6
.Q
MAIN 113
V.
SUPPLY
0 -.
11
FIL TRANS
-(D PDT
SWITCH
IS V.A C TO ENTIRE TRANSMITTER
Figure 8
COMBINED MAIN SWITCH AND
SAFETY SIGNAL
When shutting down the transmitter, throw
the main switch to neutral. If work is to be
done on the transmitter, throw the switch all
the way to "pilot," thus turning on the green
pilot lights on the panel and on each chassis, and insuring that no voltage can exist
on the primary of any transformer, even by
virtue of a short or accidental ground.
mean that the circuit is on but the lamp is
burned out or not making contact.
To enable you to touch the tank coils in
your
it is
cept
sers
transmitter with absolute assurance that
impossible for you to obtain a shock exfrom possible undischarged filter conden(see following topic for elimination of
this hazard), it is only necessary to incorporate a device similar to that of figure 8. It is
placed near the point where the main 110 -volt
leads enter the room (preferably near the door)
and in such a position as to be inaccessible
to small children. Notice that this switch breaks
both leads; switches that open just one lead
do not afford complete protection, as it is
sometimes possible to complete a primary circuit through a short or accidental ground. Breaking just one side of the line may be all right
for turning the transmitter on and off, but when
you are going to place an arm inside the transmitter, both 110 -volt leads should be broken.
When you are all through working your transmitter for the time being, simply throw the
main switch to neutral.
When you find it necessary to work on the
transmitter or change coils, throw the switch
so that the green pilots light up. These can be
ordinary 6.3-volt pilot lamps behind green
bezels or dipped in green lacquer. One should
be placed on the front panel of the transmitter;
others should be placed so as to be easily
visible when changing coils or making adjustments requiring the operator to reach inside
the transmitter.
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Safety Precautions
HANDBOOK
389
113 VOLT SUPPLY FOR
ENTIRE TRANSMITTER
AT OPERATING POSITION
TRANSMIT
STOP
USES
RECEIVE
I
SAFETY SWITCH
(SEE FIG.12)
L-t-~y-
PROTECTIVE
-CL ---121aCcH
INTERLOCKS
OVERLOAD'
CONTACTS
ORECEIVER POWER
Lt
TRANSFORMER C.T.
THERMAL
TIME-DELAY
RELAY
HIGH VOLT.
FILS. ON
,000,
STANDBY
(I
(I13 V.)
13 V. ANTENNA
CHANGEOVER
RELAY
TUNE-UP
SWITCH
INDICATOR LIGHTS
1000,
IIII17
SW.,
ALL FILAMENT TRANSFORMERS
EXCITER M.V.
TRANSFORMER
Figure
fllllll
HIGH VOLTAGE
TRANSFORMER
7
PUSH -BUTTON TRANSMITTER -CONTROL CIRCUIT
Pushing the START button either at the transmitter or at the operating position will light all
filaments and start the time -delay r e I a y in its cycle. When the c y c l e has been completed, a
touch of the TRANSMIT button will put the transmitter on the air and disable the receiver. Pushing the RECEIVE button will disable the transmitter and restore the receiver. Pushing the STOP
button will instantly drop the entire transmitter from the a -c line. If desired, a switch may be
placed in series with the lead from the RECEIVE button to the protective interlocks; opening
the switch will make it impossible for any person accidentally to put the transmitter on the air.
Various other safety provisions, such as the protective- interlock arrangement described in the
text have been incorporated.
With the circuit arrangement shown for the overload -relay contacts, it is only necessary to use
a simple normally - closed d -c relay with a variable shunt across the coil of the relay. When
the
current through the coil becomes great enough to open the normally-closed contacts the hold circuit on the plate-voltage relay will be broken and the plate voltage will be removed. If the
overload is only momentary, such as a modulation peak or a tank flashover, merely pushing the
TRANSMIT button will again put the transmitter on the air. This simple circuit provision elimi-
nates the requirement for expensive overload relays of the mechanically -latching type, but
gives excellent overload protection.
button momentarily to light the transmitter filaments and start the time -delay relay in its cycle. When the standby light comes on it is only
necessary to touch the TRANSMIT button to
put the transmitter on the air and disable the
receiver. Touching the RECEIVE button will
turn off the transmitter and restore the receiver.
After a period of operation it is only necessary
to touch the STOP button at either the transmitter or the operating position to shut down
the transmitter. This type of control arrangement is called an electrically- locking push -to-
still
transmit control system. Such systems are frequently used in industrial electronic control.
20 -3
Safety Precautions
The best way for an operator to avoid serious accidents from the high voltage supplies
of a transmitter is for him to use his head, act
only with deliberation, and not take unneces-
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RADIO
THE
Transmitter Keying and Control
388
VOLT SUPPLY FOR
ENTIRE TRANSMITTER
115
FUSES
SAFETY SWITCH
(SEE FIGS )
\St
°RECEIVER POWER
TRANSFORMER C.T.
HUSKY TOGGLE SWITCH
ON TRANSMITTER
PROTECTIVE
INTERLOCKS
THERMAL
TIME -DELAY
RELAY
O
O
TRANSMITRECEIVE SWITCH
O
HIGH VOLT.
FI
LS.
STANDS
(115V.)
O11S V. ANTENNA
CHANGEOVER
O RELAY
TUNE -UP
SWITCH
INDICATOR LIGHTS
,Qoo,
,000,
?-1-AMENT TRANSFORMERS
ALL
r1
-.000,
3V.
,Qoo)
EXCITER M.V.
TRANSFORMER
.000,
HIGH VOLTAGE
TRANSFORMER
Figure 6
TRANSMITTER CONTROL CIRCUIT
Closing S1 lights all filaments in the transmitter and starts the time -delay relay in its cycle.
When the time -delay relay has operated, closing the transmit -receive switch at the operating position will apply plate power to the transmitter and disable the receiver. A tune-up switch hos
been provided so that the exciter stages may be tuned without plate voltage on the final
amplifier.
mister on the air, has had the experience of
having to throw several switches and pull or
insert a few plugs when changing from receive
to transmit. This is one extreme in the direction of how not to control a transmitter. At the
other extreme we find systems where it is only
necessary to speak into the microphone or
touch the key to change both transmitter and
receiver over to the transmit condition. Most
amateur stations are intermediate between the
two extremes in the control provisions and use
some relatively simple system for transmitter
control.
In figure
5 is shown an arrangement which
protects mercury -vapor rectifiers against premature application of plate voltage without
resorting to a time -delay relay. No matter which
switch is thrown first, the filaments will be
turned on first and off last. However, double pole switches are required in place of the usual
single -pole switches.
When assured time delay of the proper interval and greater operating convenience are desired, a group of inexpensive a -c relays may
be incorporated into the
circuit to give a control circuit such as is shown in figure 6. This
arrangement uses a 115 -volt thermal (or motoroperated) time -delay relay and a d -p -d -t 115 volt control relay. Note that the protective
interlocks are connected in series with the
coil of the relay which applies high voltage to
the transmitter. A tune -up switch has been included so that the transmitter may be tuned up
as far as the grid circuit of the final stage is
concerned before application of high voltage
to the final amplifier. Provisions for operating an antenna- changeover relay and for cutting the plate voltage to the receiver when the
transmitter is operating have been included.
A circuit similar to that of figure 6 but incorporating push- button control of the transmitter is shown in figure 7. The circuit features
a set of START -STOP and TRANSMIT-RECEIVE buttons at the transmitter and a separate set at the operating position. The control
push buttons operate independently so that
either set may be used to control the transmitter. It is only necessary to push the START
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HANDBOOK
Transmitter Control
387
TO EXCITER POWER SUPPLIES
TO N.V.
POWER SUPPLY
C
115V.A.0
LINE
',INTERLOCKS
Jt IN TRANSMITTER
LINE
PLUGI
TRANSMITTER
GREEN
FILAMENT
PILOT
POWER CONTROL
TRANSFORMERS
RELAY
PLUG FOR
ABLE TO
VARIAC OR
115 V. TO EXCITER AND
HIGH.VOLTAGE RELAYS.
AND TO RECEIVER CONTROL AND ANTENNA
ROWERSTAT
RED
PI LOT
DUMMY PLUG FOR
STRAIGHT OPERATION
L
FILAMENT
TRANSFORMERS
TO
EXTERNAL VARIAC
OR POWERSTAT
Figure 4
CIRCUIT WITH VARIABLE -RATIO
AUTO -TRANSFORMER
When the dummy plug is inserted into the receptacle on the equipment, closing of the
power control relay will apply full voltage
to the primaries. With the cable from the
Variac or Powerstat plugged into the socket
the voltage output of the high -voltage power
supply may be varied from zero to about IS
per cent above normal.
One convenient arrangement for using a
Variac or Powerstat in conjunction with the
high -voltage transformer of a transmitter is
illustrated in figure 4. In this circuit a heavy
three -wire cable is run from a plug on the transmitter to the Variac or Powerstat. The Variac
or Powerstat then is installed so that it is accessible from the operating desk so that the
input power to the transmitter may be controlled during operation. If desired, the cable
to the Variac or Powerstat may be unplugged
from the transmitter and a dummy plug inserted
in its place. With the dummy plug in place the
transmitter will operate at normal plate voltage.
This arrangement allows the transmitter to be
wired in such a manner that an external Variac
or Powerstat may be used if desired, even
though the unit is not available at the time
that the transmitter is constructed.
Notes on the Use
of the Variac
or Powerstat
Plate voltage to the modula -
tors may be controlled at the
same time as the plate voltage to the final amplifier is
varied if the modulator stage uses beam tetrode
tubes; variation in the plate voltage on such
tubes used as modulators causes only a moderate change in the standing plate current.
Since the final amplifier plate voltage is being
controlled simultaneously with the modulator
CHANGEOVER RELAYS
52
J
Figure 5
PROTECTIVE CONTROL CIRCUIT
With this circuit arrangement either switch
may be closed first to light the heaters of
all tubes and the filament pilot light. Then
when the second switch is closed the high
voltage will be applied to the transmitter and
pilot will light. With a 30- second delay between the closing of the first switch
and the closing of the second, the rectifier
tubes will be adequately protected. Similarly,
the opening of either switch will remove
plate voltage from the rectifiers while the
heaters remain lighted.
the red
plate voltage, the conditions of impedance
match will not be seriously upset. In several
high power transmitters using this system, and
using beam -tetrode modulator tubes, it is possible to vary the plate input from about 50
watts to one kilowatt without a change other
than a slight increase in audio distortion at
the adjustment which gives the lowest power
output from the transmitter.
With triode tubes as modulators it usually
will be found necessary to vary the grid bias
at the same time that the plate voltage is
changed. This will allow the tubes to be operated at approximately the same relative point
on their operating characteristic when the plate
voltage is varied. When the modulator tubes are
operated with zero bias at full plate voltage, it
will usually be possible to reduce the modulator voltage along with the voltage on the
modulated stage, with no apparent change in
the voice quality. However, it will be necessary
to reduce the audio gain at the same time that
the plate voltage is reduced.
20 -2
Transmitter
Control Methods
Almost everyone, when getting a new trans-
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386
Transmitter Keying
TO EXCITER POWER SUPPLIES
T
s
o
o
V.A.0
LINE
115
HI -LO
POWER RELAY
POWER CONTROL RELAY
TO FILAMENT TRANSFORMERS
o
s
'Tv
o
230
v. A C
SINGLE PHASE
WITH GROUNDED
HI -LO
K2 POWER
K1
RELAY
NEUTRAL
POWER CONTROL RELAY
TRANSFORMERS
Figure
3
FULL -VOLTAGE /HALF -VOLTAGE
POWER CONTROL SYSTEMS
The circuit at (A) is for use with o 115 -volt
a -c line. Transformer T is of the standard
type having two 11S -volt primaries; these
primaries are connected in series for half voltage output when the power control relay
Kt is energized but the hi -lo relay K2 is not
operated. When both relays are energized the
full output voltage is obtained. At (B) is a
circuit for use with a standard 230 -volt residence line with grounded neutral. The two
relays control the output of the power sup-
plies the
THE
and Control
some as at (A).
primaries in parallel will deliver full output
from the plate supply. Then when the two primaries are connected in series and still operated from the 115 -volt line the output voltage
from the supply will be reduced approximately
to one half. In the case of the normal class C
amplifier, a reduction in plate voltage to one
half will reduce the power input to the stage
to one quarter.
If the transmitter is to be operated from a
230 -volt line, the usual procedure is to operate
the filaments from one side of the line, the
RADIO
low- voltage power supplies from the other side,
and the primaries of the high -voltage transformer across the whole line for full power
output. Then when reduced power output is
required, the primary of the high -voltage plate
transformer is operated from one side to center
tap rather than across the whole line. This
procedure places 115 volts across the 230 -volt
winding the same as in the case discussed in
the previous paragraph. Figure 3 illustrates
the two standard methods of power reduction
with a plate transformer having a double primary; (A) shows the connections for use with
a 115 -volt line and (B) shows the arrangement
for a 230 -volt a -c power line to the transmitter.
The full- voltage /half- voltage methods for
controlling the power input to the transmitter,
as just discussed, are subject to the limitation
that only two levels of power input (full power
and quarter power) are obtainable. In many
cases this will be found to be a limitation to
flexibility. When tuning the transmitter, the
antenna coupling network, or the antenna system itself it is desirable to be able to reduce
the power input to the final stage to a relatively low value. And it is further convenient
to be able to vary the power input continuously from this relatively low input up to the full
power capabilities of the transmitter. The use
of a variable -ratio auto -transformer in the circuit from the line to the primary of the plate
transformer will allow a continuous variation
in power input from zero to the full capability
of the transmitter.
Variable -Ratio
There are several types
Auto- Transformers
of variable -ratio auto- transformers available on the
market. Of these, the most common are the
Variac manufactured by the General Radio
Company, and the Pouerstat manufactured by
the Superior Electric Company. Both these
types of variable -ratio transformers are excellently constructed and are available in a wide
range of power capabilities. Each is capable
of controlling the line voltage from zero to
about 15 per cent above the nominal line voltage. Each manufacturer makes a single -phase
unit capable of handling an output power of
about 175 watts, one capable of about 750 to
800 watts, and a unit capable of about 1500 to
1800 watts. The maximum power- output capability of these units is available only at approximately the nominal line voltage, and must
be reduced to a maximum current limitation
when the output voltage is somewhat above or
below the input line voltage. This, however, is
not an important limitation for this type of
application since the output voltage seldom
will be raised above the line voltage, and when
the output voltage is reduced below the line
voltage the input to the transmitter is reduced
accordingly.
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Transmitter Control
HANDBOOK
not drop more than 5 volts (assuming a 117 volt line) under load and the wiring does not
overheat, the wiring is adequate to supply the
transmitter. About 600 watts total drain is the
maximum that should be drawn from a 117 -volt
lighting outlet or circuit. For greater power,
a separate pair of heavy conductors should be
run right from the meter box. For a 1 -kw. phone
transmitter the total drain is so great that a
230 -volt "split" system ordinarily will be required. Most of the newer homes are wired with
this system, as are homes utilizing electricity
for cooking and heating.
With a three -wire system, be sure there is
no fuse in the neutral wire at the fuse box. A
neutral fuse is not required if both "hot" legs
are fused, and, should a neutral fuse blow,
there is a chance that damage to the radio
transmitter will result.
If you have a high power transmitter and do
a lot of operating, it is a good idea to check
on your local power rates if you are on a
straight lighting rate. In some cities a lower
rate can be obtained (but with a higher "minimum") if electrical equipment such as an
electric heater drawing a specified amount
of current is permanently wired in. It is not
required that you use this equipment, merely
that it be permanently wired into the electrical
system. Naturally, however, there would be no
saving unless you expect to occupy the same
dwelling for a considerable length of time.
Outlet Strips
The outlet strips which have
been suggested for installation
in the baseboard or for use on the rear of a desk
are obtainable from the large electrical supply
houses. If such a house is not in the vicinity
it is probable that a local electrical contractor
can order a suitable type of strip from one of
the supply house catalogs. These strips are
quite convenient in that they are available in
varying lengths with provision for inserting
a -c line plugs throughout their length. The
a -c plugs from the various items of equipment
on the operating desk then may be inserted
in the outlet strip throughout its length. In
many cases it will be desirable to reduce the
equipment cord lengths so that they will plug
neatly into the outlet strip without an excess
to dangle behind the desk.
Contactors and
The use of power -control con tactors and relays often will
add considerably to the operating convenience of the station installation.
The most practicable arrangement usually is
to have a main a -c line switch on the front of
the transmitter to apply power to the filament
transformers and to the power control circuits.
It also will be found quite convenient to have
a single a -c line switch on the operating desk
Relays
385
to energize or cut the power from the outlet
strip on the rear of the operating desk. Through
the use of such a switch it is not necessary to
remember to switch off a large number of separate switches on each of the items of equipment on the operating desk. The alternative
arrangement, and that which is approved by the
Underwriters, is to remove the plugs from the
wall both for the transmitter and for the operating -desk outlet strip when a period of oper-
ation has been completed.
While the insertion of plugs or operation of
switches usually will be found best for applying the a -c line power to the equipment, the
changing over between transmit and receive
can best be accomplished through the use of
relays. Such a system usually involves three
relays, or three groups of relays. The relays
and their functions are: (1) power control relay
for the transmitter -applies 115 -volt line to the
primary of the high- voltage transformer and
turns on the exciter; (2) control relay for the
receiver -makes the receiver inoperative by
any one of a number of methods when closed,
also may apply power to the v.f.o. and to a
keying or a phone monitor; and (3) the antenna
changeover relay- connects the antenna to the
transmitter when the transmitter is energized
and to the receiver when the transmitter is not
operating. Several circuits illustrating the application of relays to such control arrangements
are discussed in the paragraphs to follow in
this chapter.
Controlling Transmitter
Power Output
It
is
necessary,
in
order to comply with
FCC regulations, that
transmitter power output be limited to the minimum amount necessary to sustain communication. This requirement may be met in several
ways. Many amateurs have two transmitters;
one is capable of relatively high power output
for use when calling, or when interference is
severe, and the other is capable of considerably less power output. In many cases the
lower powered transmitter acts as the exciter
for the higher powered stage when full power
output is required. But the majority of the amateurs using a high powered equipment have
some provision for reducing the plate voltage
on the high -level stages when reduced power
output is desired.
One of the most common arrangements for
obtaining two levels of power output involves
the use of a plate transformer having a double
primary for the high -voltage power supply. The
majority of the high -power plate transformers
of standard manufacture have just such a dual primary arrangement. The two primaries are
designed for use with either a 115 -volt or 230 volt line. When such a transformer is to be
operated from a 115 -volt line, operation of both
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384
__
-
FROM LINE
THE
Transmitter Keying and Control
TO OTHER
HOUSE CIRCUITS
RADIO
SHORT CORDS FROM
RECEIVER VF.O..CLOCR
FRED METER,
OUTLET STRIP.
PLAN OA
TO
PLAN pB
Figure 1
THE PLAN (A) POWER SYSTEM
A -c line power from the main fuse box in the
house Is run separotely to the receiving
equipment and to the transmitting equipment.
Separate switches and fuse blocks then are
available for the transmitters and for the
auxiliary equipment. Since the fuses in the
boxes at the operating room will be in series
with those at the main fuse box, those in the
operating room should have a lower rating
than those at the main fuse box. Then It will
always be possible to replace blown fuses
without leaving the operating room. The fuse
boxes can conveniently be located alongside
one another on the walla the operating room.
type. It is possible also that the BX cable will
have to be permanently affixed to the transmitter with the connector at the fuse -box end.
These details may be worked out in advance
with the electrical inspector for your area.
The general aspects of Plan ( B) are shown
in figure 2. The basic difference between the
two plans is that (A) represents a permanent
installation even though a degree of mobility
is allowed through the use of BX for power
leads, while plan (B) is definitely a temporary
type of installation as far as the electrical inspector is concerned. While it will be permissible in most areas to leave the transmitter
cord plugged into the outlet even though it is
turned off, the Fire Insurance Underwriters
codes will make it necessary that the cord
which runs to the group of outlets at the back
of the operating desk be removed whenever the
equipment is not actually in use.
Whether the general aspects of plans (A) or
(B) are used it will be necessary to run a number of control wires, keying and audio leads,
and an excitation cable from the operating desk
Figure 2
THE PLAN (B) POWER SYSTEM
This system is less convenient than the (A)
system, but does not require extensive rewiring of the electrical system within the
house to accommodate the arrangement. Thus
it is better for a temporary or semi-permanent
installation. In most cases it will be necessary to run an extra conduit from the main
fuse box to the outlet from which the transmitter is powered, since the standard arrangement in most houses is to run all the outlets
In one room (and sometimes all in the house)
from a single pair of fuses and leads.
to the transmitter. Control and keying wires
can best be grouped into a multiple -wire rubber covered cable between the desk and the transmitter. Such an arrangement gives a good appearance, and is particularly practical if cable
connectors are used at each end. High -level
audio at a moderate impedance level (600 ohms
or below) may be run in the same control cable
as the other leads. However, low -level audio
can best be run in a small coaxial cable. Small
coaxial cable such as RG -58 /U or RG -59/U
also is quite satisfactory and quite convenient
for the signal from the v.f.o. to the r -f stages
in the transmitter. Coaxial -cable connectors of
the UG series are quite satisfactory for the
terminations both for the v -f -o lead and for any
low -level audio cables.
Checking an
To make sure that an outlet will
Outlet with a stand the full load of the entire
transmitter, plug in an electric
heater rated at about 50 per cent
greater wattage than the power you expect to
draw from the line. If the line voltage does
Heavy Load
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CHAPTER TWENTY
Transmitter Keying and Control
20 -1
Since the usual home outlet is designed to
handle only about 600 watts maximum, the
transmitter, unless it is of relatively low power,
should be powered from another source. This
procedure is desirable in any event so that the
voltage supplied to the receiver, frequency control, and frequency monitor will be substantially constant with the transmitter on or off
the air.
So we come to two general alternative plans
with their variations. Plan (A) is the more desirable and also the most expensive since it
involves the installation of two separate lines
from the meter box to the operating position
either when the house is constructed or as an
alteration. One line, with its switch, is for the
transmitters and the other line and switch is
for receivers and auxiliary equipment. Plan
( B) is the more practicable for the average amateur, but its use requires that all cords be removed from the outlets whenever the station
is not in use in order to comply with the electrical codes.
Figure 1 shows a suggested arrangement for
carrying out Plan (A). In most cases an installation such as this will require approval of the
plans by the city or county electrical inspector.
Then the installation itself will also require
inspection after it has been completed. It will
be necessary to use approved outlet boxes at
the rear of the transmitter where the cable is
connected, and also at the operating bench
where the other BX cable connccts to the outlet strip. Also, the connectors at the rear of
the transmitter will have to be of an approved
Power Systems
It is probable that the average amateur station that has been in operation for a number
of years will have at least two transmitters
available for operation on different frequency
bands, at least two receivers or one receiver
and a converter, at least one item of monitoring or frequency measuring equipment and
probably two, a v.f.o., a speech amplifier, a
desk light, and a clock. In addition to the
above 8 or 10 items, there must be an outlet
available for a soldering iron and there should
be one or two additional outlets available for
plugging in one or two pieces of equipment
which are being worked upon.
It thus becomes obvious that 10 or 12 outlets connected to the 115-volt a -c line should
be available at the operating desk. It may be
practicable to have this number of outlets installed as an outlet strip along the baseboard
at the time a new home is being planned and
constructed. Or it might be well to install
the outlet strip on the operating desk so as
to have the flexibility of moving the operating
desk from one position to another. Alternatively, the outlet strip might be wall mounted
just below the desk top.
of all the
items of equipment, other than
transmitters, used at the operating position is totalled, you probably will
find that 350 to 600 watts will be required.
Power Drain
Per Outlet
When the power drain
383
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HANDBOOK
Deluxe Mobile Transceiver
A complete chassis -assembly mock -up should
be made up of cardboard sheets, and the
various parts laid out in order to ascertain
their final position. The tuning capacitor gang
is made up of two dual units and two single
units, with their shafts cut to length so that
the over-all depth of the gang allows room
for the p.a. plate coil and associated padding
capacitors.
done this type of assembly and wiring as a
vocation will find this style of construction
interesting and a challenge to his ingenuity.
The beauty of the final equipment is well
worth the time and study it takes to design
and lay out a unit of this order of complexity.
Transceiver
Alignment
and Test
Transceiver
The under -chassis wiring may
Wiring
be observed in figure 38. All
power wiring is laced to form
a harness that runs about the chassis in a
square loop centered about the coil assembly.
Small components are mounted directly to tube
socket pins, to lug terminal strips, or to small
phenolic terminal boards. Ground connections
are made to lugs placed beneath socket retaining bolts.
The r.f. components of the receiver occupy
the center portion of the chassis. Small inter stage shields made of durai separate the r.f.,
mixer, and oscillator stages, and an additional
shield plate covers the bottom of the 6AH6
v.f.o. compartment. To the rear of this compartment are the driver stages of the transmitter section.
A wiring harness of the type used in this
transceiver may best be made up external to
the unit. A layout of the harness and the
terminations of the various wires is sketched
full -size on a large board and the wires are
then laid out on the board in their proper positions, cut to length, and laced. The completed
harness is then dropped into the equipment
and the terminations made. An amateur experienced in equipment construction, or who has
TRANSCEIVER
Amplifier tuning and
loading
controls are
mounted on rear of the
cabinet. Below (left to
right) are: antenna receptacle, power receptacle, speaker receptacle,
and 5 -meter zero -set po-
tentiometer. Additional
ventilation is provided by
rows of holes across rear
of cabinet.
11
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When the transceiver is completed, all wiring should be
checked and "rung out" to preclude the possibility of wiring
errors or accidental grounds. The tubes are
now placed in the unit, and the various tuned
circuits adjusted to their approximate operating range by means of a grid-dip oscillator.
The transmitter and modulator tubes are removed, and the receiver section is aligned in
the following manner: The first step is to
align the low frequency i.f. strip. A low level
modulated 260 kc. signal is injected into the
plate circuit of the 6BE6 second mixer and
transformers T.., T3, and T4 are adjusted for
maximum receiver output. Next, oscillator coil
L of the 6BE6 stage is adjusted for maximum #1 grid current and a 4.26 Mc. signal
is fed to the input circuit of the 6BA7 first
mixer. Transformer Tt is adjusted for
maximum signal strength.
A 29 Mc. signal is now applied to the
antenna circuit of the receiver, and the main
tuning dial is adjusted to this approximate
setting. Coil L3 and capacitor Ca of the master
oscillator are adjusted until the test signal is
heard. The tuned circuit of the oscillator is
aligned to cover the span of 23,740- 25,440
kc., with equal leeway on each end of the
range. The test signal is now placed on 29.5
Mc. and the padding capacitors of the r.f. and
Figure 39
REAR VIEW OF
563
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tt
564
THE RADIO
Receivers and Transceivers
mixer stage are adjusted for maximum signal.
The signal is next shifted to 28.5 Mc. and
the variable slugs of the r.f. and mixer coils
are adjusted in turn. This process is repeated
until the tuning range of the receiver is correct,
and the r.f. stages track properly across the dial.
The transmitter section may now be aligned.
The tubes are inserted in their sockets and
relay RY1 is activated. The screen power lead
to the 6146 is temporarily opened to disable
that stage. Once again, the transceiver is tuned
to 29.5 Mc. and the two padding capacitors of
the 6CL6 buffer stages are adjusted for maximum grid drive to the 6146 stage. (Note:
Grid current to the 6146 should be held to
less than 4 ma. at all times) . The dial is now
returned to 28.5 Mc. and the variable slugs
of the buffer circuits are adjusted for maximum grid drive. The adjustments are repeated
until reasonably constant grid drive occurs
across the tuning range. The buffer stage and
power amplifier are neutralized and screen
voltage is applied to the 6146 tube. The
transmitter frequency is set to 29.0 Mc. and
the amplifier is tuned and loaded by means
of the controls on the rear of the cabinet
(figure 39) . The frequency of the transceiver
is shifted to 29.5 Mc. and (without adjusting
the loading capacitor) resonance is again reestablished with the rear tuning capacitor,
Now, the frequency is shifted to 28.5 Mc.
and auxiliary capacitor C14 is adjusted for
resonance. This sequence of adjustment is repeated until proper resonance and loading
occurs across the dial. Resonant plate current
should be approximately 110 milliamperes
and grid current should be 2 to 3 milliamperes.
Modulator resting plate current is 25 milliamperes, rising to about 80 milliamperes
under full modulation.
C.
The transmitter may be bench -tested with
an a.c. power supply and a dummy load before
it is placed in the automobile. Car mounting
is accomplished by means of two heavy aluminum rails bolted to the top of the transceiver case which slide into suitable clamps
affixed under the dash of the automobile as
shown in figure 31. A transistor -type power
supply or a dynamotor may be used. 250 volts
at about 150 milliamperes, and 500 -600 volts
at 200 milliamperes are required for operation
of the transceiver.
27 -7
A Deluxe
Receiver for the
DX Operator
The need exists for a high performance
receiver, suitable for s.s.b., a.m., and c.w.
operation that can be built in the home workshop at a modest price. The receiver should
have a high order of stability and sensitivity
and must have sufficient dynamic range to
protect it against excessive cross- modulation
caused by strong nearby signals. In addition,
it should be possible to build the receiver
without the use of special metal- handling tools.
The receiver described in this section was
designed to fill this need. It is a double conversion superheterodyne, employing crystal control in the first conversion stage and a tunable
low frequency i.f. and mixer. This configuration provides maximum stability and permits
the use of a dial calibrated directly in
frequency.
Collins mechanical filters and a Q- multiplier are used in the 455 kc. second intermediate frequency amplifier to provide the
ultimate in selectivity and rejection and a
product detector is employed for c.w. and
Figure 40
FRONT VIEW OF DELUXE AMATEUR
COMMUNICATION RECEIVER
band receiver covers 80 -10 meters, with
extra bond for 15 Mc. reception of WWV
standard frequency signals. Collins mechanical
filters provide ultimate in selectivity for s.s.b.
a.m. phone, and c.w. The receiver employs a
crystal controlled first conversion oscillator
for high -order stability and "hang- a.g.c." for
improved sideband reception. A simplified
product detector is used for s.s.b. and c.w.
operation. The precision dial can be read to
one or two kilocycles on all bands. Room is
provided above main dial for inclusion of
v.h.f. converters for 2 and 6 meter operation,
if desired.
Six
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HANDBOOK
V2
V1
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DX Communication Receiver
9
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TUNING
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Ve
AM-CW-55B
2ND AUDIO
VIS
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VOLT. REGULATOR
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12AU7
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Figure 41
BLOCK DIAGRAM OF DELUXE AMATEUR COMMUNICATION RECEIVER
reception. An automatic gain control
circuit (a.g.c.) is provided for sideband, and
auxiliary equipment includes an S -meter and
100 kc. crystal calibrator. Reception of the 15
Mc. Standard Frequency (WWV) signal is
incorporated for receiver calibration purposes.
Construction is simplified by making the
receiver in modules that may be built and
tested one at a time.
s.s.b.
The Receiver
diagram of the receiver circuit is shown in figure
41. Fourteen tubes are used,
plus a voltage regulator. The power supply
utilizes semiconductors to reduce heating
effects.
The R.F. Section. The receiver covers the
amateur bands between 10 and 80 meters,
with an extra bandswitch position for "spot"
reception of WWV at 15 Mc. The r.f. stage
employs a 6DC6 semi -remote cutoff pentode
to provide maximum freedom from crosstalk
and front -end overload. A triode -connected
6AH6 serves as a low noise mixer stage, with
local oscillator injection on the control grid.
The first conversion oscillator is crystal controlled using a 6BJ6 in a "hot cathode" circuit
operating on the low frequency side of the
received signal.
Circuit
A block
Receiver tuning is accomplished at the first
intermediate frequency range of 2.4 -2.9 megacycles. Each tuning range thus covers 500
kilocycles. Any 500 kc. segment of the 10
meter band may be utilized by the proper
choice of the conversion crystal. The tunable
portion of the receiver consists of a 6BJ6 i.f.
amplifier, and a 6BE6 second mixer stage. The
oscillator portion of the 6BE6 tube tunes the
region 1.945 -2.455 Mcs. to provide a 455 kc.
intermediate frequency. Both oscillators are
voltage regulated for maximum frequency
stability.
The I.F. Section. Two i.f. stages are employed
to provide sufficient receiver gain. The first
stage uses a 6AH6 which directly follows the
mechanical filters and the Q- multiplier circuit.
The filters allow a choice of 0.5 kc. passband
for c.w., or a 3.0 kc. passband for sideband.
A.m. reception may be done by listening to
one of the two sidebands, or a 6.0 kc. bandwidth filter may be substituted for the 3.0 kc.
unit. The Q- multiplier places a rejection
"notch" at any point in the filter passband to
eliminate heterodyne interference. The depth
of notch can be adjusted by an auxiliary
control.
The over -all gain of the receiver is set by
adjusting the "r.f. gain" control which fixes
the operating bias on the low frequency i.f.
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Figure 42
page)
(See opposite
-50
Aµfd. National UM -50 or equivalent
UM -100 or equivalent
C}
F. Johnson SMB11, with 240 µµtd. silver mica
shunted across each section
Ci -22 gµl& silver mica capacitor with 7 -3S µµI&
ceramic trimmer connected in parallel
C, -120 µµtd. silver mica capacitor with 7-35 µµfd.
ceramic trimmer connected in parallel
RFC
mh. J. W. Miller Co. 1:J300-1000
S,4,n,e,n- Centralab PA -305 assembly with 6 -inch
shalt and six Centralab PA -17 ceramic sections
(60 degree index)
assembly with 4 -inch
SSA,n,e --- Centralab PA -301
shaft and two Centralab PA -0 ceramic sections
S3- Corner Plates of C, bent to short out filter
T1, T2
W. Miller Co. #B -727RF coil with S -27
shield
Lr
W. Miller Co. re-727C coil with S -27 shield
X, Crystals -International Crystal Mfg. Co.
Dial- Eddystone. Available from British Radio Electronics, Ltd., 1833 Jefferson Place, N.W., Washington 6, D. C.
All bypass capacitors are .01 ;lid., disc ceramic, 600
volt. High frequency oscillator capacitors are silver
mica
Mechanical filters- Collins Radio Co., 455 kc., style K
C1-C4
C5-100 µofd. National
-E.
-1
-J.
-J.
stages and also on the tunable i.f, stage. The
front end of the receiver operates at maximum
sensitivity and gain at all times in order to
override the inherent tube noise level of the
various mixer stages.
The Detector and Audio Section. A 6BE6
mixer tube is employed as a hybrid detector.
For sideband and c.w. operation, it functions
as a product detector, with injection on the
#1 grid from the beat oscillator and signal
injection on the #3 grid. For a.m. service,
the beat oscillator is disabled, and the signal
is switched to the #1 grid. Thus one tube
serves two functions, and does both of them
well. The beat oscillator is a 6BJ6, with variable injection taken from the plate circuit.
The oscillator frequency may be moved across
the passband of the i.f. system to provide a
choice of upper or lower sideband reception,
as desired.
The automatic gain control system employs
a separate 6BJ6 i.f. amplifier stage driving a
simple "hang- a.g.c." system of the type described by W1DX in the January, 1957 issue
of QST magazine. The 6BJ6 stage isolates the
b.f.o. from the a.g.c. system and prevents
oscillator voltage from leaking into the a.g.c.
circuit. The latter circuit is especially designed
for s.s.b. and c.w. reception. It has a very
rapid response that prevents receiver overload
567
on a syllabic burst of s.s.b., instantly reducing
receiver gain to prevent overloading. The gain
reduction remains in effect as long as the
signal is in evidence, then "hangs" on for
about 0.5 second after the removal of the
signal. This sequence of action reduces to a
minimum the usual "thump" that occurs at
the start of a syllable and removes the "rush"
of background noise at the end of a syllable
that occurs with a conventional a.v.c. system.
A triple diode 6BC7 and one -half a 12AU7
double triode comprise the complete "hanga.g.c." system. The double diode system following transformer T., and the 470K/0.01
µfd. R -C network determine the "on" time
of the "attack" system, permitting the 0.1 µfd.
a.g.c. capacitor to charge up in a relatively
quick time. The capacitor remains charged, as
the 12AU7 triode is cut off by this action,
and there remains no discharge path to ground
in the a.g.c. circuit, even when the voltage
across the "attack" R -C network is removed.
The time constant of the "release" network
is considerably longer, and after a predetermined period, the a.g.c. voltage across this
network decays sufficiently to permit the triode
section to conduct and discharge the a.g.c. line
capacitor. The proper ratio of voltages in the
two R -C circuits can easily be established by
proper adjustment of transformer T. A slight
degree of delayed a.g.c. action is provided by
applying fixed bias to the "attack" diode to
prevent the circuit from being tripped by
back,.grc und noise or weak signals.
The S- Meter, Audio System and Power Supply.
The S -meter circuit is a simple vacuum tube
voltmeter that compares the a.g.c. voltage
against a fixed reference voltage. The circuit
is balanced for a meter null with no signal
input to the receiver, and a.g.c. voltage unbalances the circuit causing a reading on the
meter placed in the bridge of the circuit. The
meter may be used for all modes of reception,
providing usable readings on c.w. signals as
well as sideband or a.m.
A single 6AK6 provides sufficient audio
for earphone reception, or to drive a speaker
to good room volume. Ignition and other
pulse -type noise is effectively reduced by
means of a peak noise clipper made up of
two inexpensive semiconductor diodes.
The power supply is a voltage doubler type
utilizing inexpensive silicon rectifiers. High
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www.americanradiohistory.com
DX Communication Receiver
569
voltage is regulated by an 0A2 for the entire
receiver, and standby is accomplished by breaking the B -plus line from the supply. Three
separate filament windings on the power
transformer provide sufficient capacity to
power all the tubes. The use of 150 milliampere filament tubes wherever possible reduces the filament drain considerably. The
whole receiver runs reasonably cool because
of the low plate voltage and choice of low
filament power tubes, achieving a high order
of thermal stability in a short period of time.
Receiver
Construction
Figure 44
TUNABLE I.F. SECTION OF RECEIVER
The tunable i.f. section cf the receiver is
built upon a 3" x 5" x 7" aluminum chassis.
Input and output connections are made via
"phono- type" coaxial fittings and lengths of
RG -58 U coaxial line. Tube in foreground is
68E6 mixer (Vs), and tube in the rear is
6816 tunable i.f. (V4). Ceramic padding capacitors C, (two) and Cv are mounted at right
of chassis, with the three Li. coils atop the
A receiver such as this is a
complex device and its construction should only be under-
taken by a person familiar with receiving
equipment and who has built equipment of
this category before.
The receiver is built upon an aluminum
chassis measuring 1531" x 11" x 3" in size,
and is contained within a ventilated cabinet
measuring 16" x 111'4" x 91 ') ". The tunable
i.f. system is built as a separate unit on an
aluminum chassis -box measuring 3" x 5" x
7" (figures 44 and 45). The mechanical
filter assembly is also built as a separate unit
in an aluminum box measuring 2" x 3" x
chassis.
Figure 45
UNDER -CHASSIS
VIEW OF TUNABLE
I.F. SECTION OF
RECEIVER
Tunable i.f. stage is isolated from second mixer
by shield partition across
middle of chassis box.
Mixer and oscillator sections of Vs are separated
by a small partition.
Tuning
capacitors are
mounted to the shield
partitions and are driven
through metal shaft couplings. Power receptacle is
at rear of chassis. Complete assembly is fastened to main chassis by
six sheet metal screws.
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570
THE RADIO
Receivers and Transceivers
51/4" ( figure 47) . The b.f.o. assembly is built
within an aluminum box measuring 11/2"" x
2" x 23/ " (figure 48). The remainder of the
receiver is built upon the main chassis.
No receiver is better than its tuning dial,
so the very excellent Eddystone geared slide
rule dial is used. The dial is centered hori-
zontally on the panel and vertical placement
is adjusted so that the drive engages the shaft
of the variable tuning capacitors of the tunable
i.f. system.
Figure 46
UNDER -CHASSIS VIEW OF
COMMUNICATIONS RECEIVER
The receiver is built in sections which may be
checked out one at a time for sake of sim-
plicity. Crystal controlled r.f. section is at
left, with coil slugs projecting from front and
back of assembly. Conversion crystals are
mounted in holders on front partition. Near
center of chassis is box containing mechanical
filters and switch (figure 47). At right is partition holding Q- multiplier coil and potentiometer, with auxiliary notch control located on
the panel. The product detector switch is
driven off- center by two flexible couplings.
Power supply, diode rectifiers and filter section are at lower right, with audio stages
across bottom of chassis.
The chassis, panel, and tunable i.f. chassis
should be assembled and studied before any
chassis holes are drilled. The dial cut -out
should next be made, making sure of alignment of the dial with the variable capacitors.
Placement of the remainder of the components
is not at all critical.
The Tunable I.P. System. It is best to construct this item first, as it determines dial
position and placement of other major parts.
A close -up of this assembly is shown in figures
44 and 45. The three variable capacitors are
ganged by means of brass shaft bushings. The
first capacitor is mounted to the front wall of
the chassis -box, and the other two are placed
on aluminum interior partitions. The 6BE6
mixer tube is mounted to the front with the
6BJ6 at the rear. Power connections are made
to a miniature connector on the rear of the
chassis, and input and output terminations are
made through "phono- type" coaxial connectors
and short lengths of RG -58/U coaxial line.
The Mechanical Filter Assembly. A partition
separates the input and output circuits of the
filter assembly, as shown in figure 47. Bulk-
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HANDBOOK
DX Communication Receiver
head mounting filters are used to achieve a
maximum degree of isolation across the filter.
The individual segments of the selectivity
switch (S_) are mounted in each compartment, with the switch mechanism passing
through the bulkhead. A spring wiping contact
is made for the rotor arm of the switch,
grounding it at the center bulkhead to prevent a leakage path around the filter from
being formed. Input and output terminations
are made via "phono- type" coaxial fittings
and RG -58/U coaxial line.
The input and output circuits of the filters
must be tuned to frequency. This is accomplished by a 50 µµtd. variable padding capaFigure 47
INTERIOR VIEW OF MECHANICAL
FILTER ASSEMBLY BOX
Bulkhead mounting mechanical filters are
mounted to interior partition which isolates
input and output sections. Drive shaft of selectivity switch is grounded at point it passes
through partition by a wiping spring to achieve
maximum circuit isolation. Input and output
tuning capacitors of filters are made up of
SO ppld. variable ceramic trimmers connected
in parallel with 75 µpfd. silver mica capacitors.
Trimmers are adjusted for maximum signal
response, in same manner as i.f. transformer
capacitors.
11
571
citor placed across each circuit and adjustable
from the bottom of the receiver.
The R.T. Assembly. The r.f. assembly is constructed within the main chassis as shown in
figure 50. The sockets for the 6DC6 r.f. stage,
the 6AH6 mixer, and the 6BJ6 oscillator are
mounted on the main chassis and the associated coils, tuning capacitors, and bandswitch
are mounted to four vertical partitions fixed
beneath the chassis. Slug -tuned coils are used
for all circuits and are mounted in a horizontal position about the bandswitch. The r.f.
and mixer coils can be aligned by means of a
"TV- type" screwdriver thrust through holes in
the rear of the chassis, while the r.f. coils are
adjusted from the front of the assembly by
means of a short screwdriver. The partitions
are mounted so that a space of 2" exists between them, and the associated tube socket
falls in the center of each space. The switch
assembly passes through the partitions and, in
fact, holds them in position by virtue of the
switch arms and spacers. The individual switch
segments are placed so that they are near the
end of each coil. This results in a very compact assembly having extremely short leads to
all coils. The coils are staggered about the circumference of a circle so that both the r.f.
and mixer slugs can be reached from the rear
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572
THE RADIO
Receivers and Transceivers
without interference between the coils.
The four partition plates are cut from
1/32 -inch aluminum stock, and follow the
layout of figure 51. They are not notched at
first. Rather, a cardboard template is cut out
and marked for drilling as shown. Then all
four partitions are clamped together and
drilled along with the template. Corner notches
are now cut and all edges filed so that all four
partitions are as identical in size and shape as
possible. Only the holes shown in figure 51
are common to all pieces. The front and rear
partitions have other holes
i.e., crystal
sockets, antenna input, power lead holes, etc.
These may be drilled during layout and assembly of the unit as required. The 1/2-inch
flanges are then bent over, taking care to bend
the front and rear pieces in the proper
direction.
-
The coils should be wound to the data of
figure 49, before the unit is assembled. Only
three coils are used in the oscillator section
as an r.f. choke is employed on the 80 and
40 meter bands. The 14 Mc. coils are jumpered across the switch and used for the
WWV position on 15 Mcs. All coils should
be wired to the bandswitch before the tuning
Figure 48
REAR VIEW OF RECEIVER
Placement of major parts may be seen in
this view. B.f.o. components and tube are
mounted in small aluminum box next to the
front panel (left), with S -meter above main
tuning dial. At right on panel is standby control switch, with noise limiter switch beneath
it. Power transformer is at left rear of chassis.
On rear apron of chassis are placed !1. to r.):
115 volt power receptacle,
utility socket,
break -in gain control, S -meter adjust potentiometer, speaker terminals, and coaxial antenna receptacle. At extreme right are pass through holes to permit alignment of high
frequency r.f. coils.
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HANDBOOK
DX Communication Receiver
573
Coil Table
Figure 49
Band
L1
3/16"
1.
closewound
6t #24e
40
3/16"
3/16"
1.
3/16"
closewound
closewound
33t #24e
12t #24e
closewound
L5
RFC
RFC
16t #24e
8t #24e
40t #30e
spaced length
closewound
closewound
(11600 kc.)
14t #24e
7t #24e
23t #24e
spaced as above
closewound
spaced to cover
form
(18,600 kc.)
3t #24e
10
20t #30e
of form
3-1/2t #24e
3/16" 1.
15
55t #30e
closewound
1.
4t #34e
20
L3
Ls, L4
9t #24e
80
1.
12t #24e
6t #24e
16t #24e
spaced as above
closewound
spaced to cover
form
(25,600 kc.)
All coils wound on XR -SO forms.
Ls, L. wound first -then a loyer of 1/2" Scotch No. 33 tape wound on cold ends of L2, L4 coils
and LI, L3 primaries wound over tape. Small strip of tope plus coil cement secures the free
ends of LI, L3.
80 M coils L2, L4 have SOµ,.fd. padders soldered across terminals.
capacitors are finally mounted in place. The
last step is to use the unit as a template to
mark the clearance holes on the rear of the
chassis, which are drilled before the unit is
finally installed in the chassis.
Receiver Wiring. The remainder of the receiver wiring is simple and straightforward.
The sideband -a.m. switch (S4) is offset from
the panel hole to clear the Q- multiplier coil
(Ls) mounted on an L- shaped bracket beneath the chassis (figure 46) The audio low pass filter coil (L9) is placed between the
6BE6 detector and 12AU7 audio socket. Long
runs of a.c. leads are done in shielded wire, as
are audio leads.
.
The receiver may be aligned in
sections. The first step is to
align the i.f. system and beat
oscillator. Next, the tunable i.f. stages should
be aligned and tracked. Finally, the r.f. sections are properly tuned.
The i.f. system should be aligned to the
center of the passband of the narrowest -bandwidth mechanical filter. In the case of the
500 cycle filter, the center frequency must be
455.0 kilocycles with very little tolerance. The
Receiver
Alignment
system may be roughly aligned with the aid
of an external signal generator coupled into
the #3 grid of the 6BE6 second mixer. A
455 kc. signal of low amplitude is injected
into the input circuit and the tuning capacitors across the filter terminals, plus transformers T3 and T4 are adjusted for maximum
response. The Q-multiplier should be out of
the circuit for this test (switch S3 closed).
Care should be taken not to overload the i.f.
system during alignment, so a relatively weak
signal should be used for this portion of the
adjustment. A.g.c. transformer T5 should then
be adjusted to provide the proper "attack" and
"release" time for the gain control circuit.
Finally, the slug of the b.f.o. coil (L10) is set
to place the beat oscillator signal at the center
of the i.f. passband with the b.f.o. panel
control set at mid- scale.
The signal generator is now shifted to the
input circuit of the 6BJ6 tunable i.f. stage.
The main tuning dial is set at 500 (minimum
circuit capacitance). The generator is adjusted to 2.90 Mc., and padding capacitor Cy
of the oscillator section is adjusted for signal
response. 1.f. and mixer padders C. are then
tuned for maximum signal. At a dial reading
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574
THE RADIO
Receivers and Transceivers
Figure 50
UNDER -CHASSIS
VIEW OF R.F.
COIL ASSEMBLY
The high frequency coils
are placed in a circle
about
the
bandswitch
(figure
51). Coils and
capacitors are mounted
on four shield partitions
which are located between the tube sockets.
R.f. stage socket (V1) is
at rear of chassis, with
mixer socket (V2) in center, and crystal oscillator
socket (V3) nearest the
panel. Oscillator crystals
are mounted on front
partition. Entire assembly is shielded by aluminum cover plate.
e
of zero (maximum circuit capacitance), the
tunable stages should resonate at 2.40 Mc.
Attention should now be given to the front end stages. It is a good idea and a time saver
to peak circuits L1 -L2 and L3 -L4 to the proper
frequency with the aid of a grid -dip oscillator.
Coil L; is adjusted for proper crystal oscillator
operation, which may be monitored in a
nearby receiver. The signal generator is now
set to the center frequency of the 500 kilocycle band in use and a moderate signal is
injected into the antenna circuit of the receiver.
The main tuning dial is adjusted to receive
the signal, and the r.f. and mixer coils are
peaked for maximum response with the r.f.
tuning capacitors set at mid -scale.
Once alignment has been completed, the
operator should familiarize himself with receiver operation. The last step is to adjust
the "break -in" gain control so that the receiver
may be used to monitor c.w. transmissions. The
Figure 51
R.F. ASSEMBLY PLATES
Four assembly plates are required, as shown.
Each plate is drilled as necessary for mounting
of small components, etc.
PELATIvE POSiT,ON
WHEN A5SEMBLED
OF
TUBES
OR
UM-50
-50 FORM OUTLINE
OuTLINE
12
1
R
TB
B_
TO
--
ITCH
STATOR
TAB
L
www.americanradiohistory.com
-6
Di
NOTE OWL
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ROTOR
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SPACED EVERT 50. ON
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SCREW BOA 5 121
2
HANDBOOK
short across the control circuit is removed
from the utility socket on the rear apron and
the control adjusted for the desired standby
sensitivity level. The control may be shorted
out by an external switch or relay during
periods of reception.
Ti
6. 3 V., 4. 2 5
I.F.
A ., T O
(CREENLEADS)
Figure 52
63V,4.0
SCHEMATIC, POWER SUPPLY FOR
RECEIVER
-4.5 H at 200 ma. Stancor C-1411
S,
A -2 pole, 3 position rotary switch
SRI, -200 ma. rectifier. Sarkes- Tarzian
A.,TO
I
AUDIO
U D O
I
R. F.
(BROWN LEADS)
6.3
V., 2 OA TO TUNABLE
(YELLOW LEADS) L.
L11
SR,
,
2
575
DX Communication Receiver
M-
500 with dual mounting kit
T7-117 volts at 200 ma. Three 6.3 volt
windings at 2.0, 4.0, and 4.23 amperes,
respectively. Stancor P -8158
r
STa
OFF
sreY.
oN-0
a+
350
+ 40
5W
www.americanradiohistory.com
I.
200
E
&
E
TECHNI -SHEET
CONVERSION TABLE
MICRO
=
-
(µ) ONE -MILLIONTH
MILLI = (m) ONE -THOUSANDTH
TO CHANGE
FROM
UNITS
MICRO -UNITS
MILLI -UNITS
KILO -UNITS
KILO
(K) ONE THOUSAND
(M) ONE MILLION
MEGA
TO
OPERATOR
MICRO -UNITS
MILLI -UNITS
X 1,000,000
X 1,000
KILO -UNITS
MEGA -UNITS
±
±
MILLI -UNITS
UNITS
MICRO-UNITS
UNITS
MEGA -UNITS
UNITS
MEGA -UNITS
UNITS OF MEASUREMENT
KILO -UNITS
UNITS
-
±
X
X
1,000
or X
1,000,000 or X
or
108
or
103
10 -3
10 -8
1,000
or X 10 -3
1,000,000 or X 10 -6
X 1,000
or
=
1,000
or
-
1,000
or
X 1,000
or
X
X
103
X
X
10 -3
X 1,000
or X
1,000,000
X
or X
www.americanradiohistory.com
10 -3
103
103
108
CHAPTER TWENTY -EIGHT
Low
Power
Transmitters and Exciters
The transmitter is the "heart" of the amateur station. Various forms of amplifiers and
power supplies may be used in conjunction
with basic exciters to form transmitters which
will fit almost any requirement. Several different types of transmitting equipment designed to meet a wide range of needs are outlined in this chapter. A simple transistorized
transmitter for 50 Mc. is described. This unit
is a good introductory project for the amateur
to "cut his teeth on" relative to the field of
transistors. Also shown is a complete, TVIproof, medium- powered all -band phone and
c.w. transmitter. A "W9TO" electronic keyer
is illustrated, together with newly -developed
"Strip Line" circuits which are applicable to
the v.h.f. spectrum. For the amateur who is
interested in the construction phase of his
hobby, these units should offer interesting
ideas which might well fit in with the design
of his basic transmitting equipment.
Figure 1.
A POCKET -SIZE
50 MC. TRANSISTORIZED PHONE
TRANSMITTER.
Capable of 100 milliwatts
input,
this
"collector
modulated" six meter
phone transmitter will
provide amazing results
when used with a good
antenna system. The complete unit may be held
in the palm of the hand.
Panel controls are 11. to
r.): crystal oscillator tuning (top) and audio gain
control (bottom), multi meter, amplifier tuning
(top) and loading (bottom). Switch on left is
the multi -meter switch,
with power switch at
right. Microphone plug
is
centered between
switches.
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578
THE RADIO
Low Power Transmitters
RCA
RCA
2N3B4
2N384
5O AK
X1
(PNP)
L2
(P)(P)
LI
SO R
ADJUST
!/AS
t
OSC
w--AMP
2 N44
(PNP)
MIC
ALL RESISTORS 1/2 WATT.
15V.
-ISV
AUX.
Figure 2.
SCHEMATIC, 50 MC. TRANSISTORIZED TRANSMITTER.
L1-6
turns =18 wire, 58 inch diameter, S/8
inch long. (B8W miniductor
3007.) Top
three turns from transistor end
L2, L3 -Make both coils from a single piece
of 88W miniductor =3007. Use nine turns.
Cut coil between sixth and seventh turn,
making two coils having six and two turns,
28 -1
A Transistorized
50 Mc. Transmitter
and Power Supply
The simple 50 Mc. transistorized transmitter shown in this section makes an interesting project for the amateur who wishes to
familiarize himself with high frequency transistors. Capable of 100 milliwatts input, this
little phone transmitter will give a good
account of itself when it is used in conjunction
with a beam antenna. It may be run from
batteries or from a regulated a.c. power supply.
Circuit
The transmitter circuit utilizing inexpensive PNP -type
transistors is shown in figure
2. The oscillator is crystal controlled, employDescription
respectively, separated by a distance
one turn
M
-10 ma., d.c., 11/4" square meter
-0
T1- Transistor
of
transformer, SK to 80K. Thor -
darson TR -13
T2,
Ta-Transistor transformer,
10K to 2K.
Triad TY -56X
ing a 2N384 in conjunction with a 50 Mc.
third -overtone crystal connected between
collector and base of the drift transistor. Operating bias level is adjusted by a variable
potentiometer. The low impedance base of the
2N384 amplifier is tapped on the oscillator
coil to achieve a match to the higher impedance collector circuit of the oscillator. The
amplifier collector output circuit is inductively
coupled to the antenna. It may be seen that
this configuration bears a close similarity to a
vacuum tube circuit in that the emitter of the
transistor resembles the cathode of the tube.
The base may be compared to the grid, and the
collector to the plate.
A two stage modulator section provides
sufficient gain to operate a dynamic micro-
www.americanradiohistory.com
HANDBOOK
50 Mc. Transmitter
phone. The audio stages are tranformer
coupled and base driven. A 1N34 diode is
used as a high level positive peak loading
device to prevent peak clipping at high modulation levels. Positive peak clipping is employed since the collector supply voltage is
negative with respect to ground. A simple
metering system permits the operator to moniFigure 3.
REAR VIEW OF
TRANSISTORIZED TRANSMITTER.
The two r.f. transistors are mounted in sockets
on L- shaped bracket at the center of the
chassis. Directly below them is the oscillator
bias -potentiometer. Across the rear edge of
the chassis are the audio stages, with the
power terminals on the rear apron of the
chassis. Relative size of transmitter and com-
ponents may be judged from comparison with
standard coaxial receptacle at left of chassis.
Oscillator stage is at right, with amplifier
at left.
579
tor the collector current of the r.f. stages.
The positive terminal of the power supply
is at "ground," or chassis potential. If NPNtype transistors are substituted for the specified units, battery polarity must be reversed.
Transmitter
Construction
The complete transmitter is
built upon a small aluminum
chassis measuring
x 31/2"
x 1" in size. The front panel measures 6" x
4 ". The two r.f, transistor sockets and ri.
components are mounted on an L- shaped aluminum bracket centered on the chassis, measuring 2" high by 21/4" long. The right -angle
portion of the bracket holding the crystal
socket is 11/2" high by 1" wide. Miniature
transistor sockets are mounted in the top
corners of the bracket, with the oscillator bias
control centered beneath them (figure 3) .
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580
THE RADIO
Low Power Transmitters
The transistorized audio section is placed
across the rear of the chassis. Transformer
leads pass through small rubber grommets to
the under -chassis area. At one end of the
chassis is an aluminum bracket holding the
coaxial antenna receptacle. Small components
are mounted under the chassis on phenolic
terminal strips. Transmitter wiring is straightforward, and is done with #22 insulated wire.
Coil data is given in figure 2.
Shown in figures 5 and 6 is a simple voltage regulated power supply that provides 18
volts at 100 milliamperes. A 2N561 power
transistor is used as a series regulator, with a
2N44 serving as a regulator driver stage. The
control element is a Zener diode delivering a
constant source of 14.7 volts, which is used to
set the output voltage. As the transmitter is
operating near maximum transistor voltage
values, it is important that the power supply
Figure 4.
UNDER -CHASSIS VIEW OF TRANSMITTER.
Miniature components are mounted on phenolic terminal strips beneath the chassis.
"Clipping" diode is at right, behind slide
switch. Audio leads are run in shielded wire.
voltage remain constant under varying loads.
A voltage surge could possibly damage the
transistors in the transmitter at this relatively
high operating potential.
The power supply is built upon an aluminum chassis measuring 51/2" x 31/2" x 1 ".
The 2N561 power transistor must be insulated from the chassis by means of mica shims
or an anodized plate, as the collector element
is bonded to the case of the unit. The power
supply may be tested by placing a 350 ohm,
10 watt resistor across the output. 18 volts
should be developed across the resistor.
Transmitter
Adjustment
and Tune -up
When the transmitter wiring
is
completed, it should be carefully checked, especially in the
area of the transistor sockets.
Insert the r.f. transistors and crystal in their
sockets and turn the oscillator bias potentiometer to maximum resistance. Place the meter
switch in the oscillator position. Use a 52 ohm,
1 -watt composition resistor across the antenna
receptacle as a dummy load for these tests.
Turn the transmitter on and adjust the oscillator tuning capacitor for oscillation ( jump in
collector current) as noted on the meter. Ad-
www.americanradiohistory.com
HANDBOOK
TI
SRI
200 -Watt Transmitter
-
Figure S.
SCHEMATIC,
VOLTAGE REGULATED POWER SUPPLY.
B -115 volt neon lamp in holder
5R,
Silicon rectifier, 400 v. p.i.v., S00 ma.
Sarkes -Tarzian CM -500
-4-
T,-
Filament transformer. 26.8 volts at 1 a.
Triad F -40X
Z, -Zener diode, 15 volts, 1/2 watts, Motorola 1.5M15Z (10% tolerance)
just the bias potentiometer for about 5 milliamperes oscillator current. Now, place the
meter switch in the amplifier position and
adjust the oscillator tuning capacitor for maximum meter reading. Adjust the amplifier tuning capacitor for a meter dip. Finally, adjust
the antenna loading until the meter indicates
about 6 milliamperes, re- resonating the circuit
with the collector tuning capacitor. A field
strength meter is helpful for the initial
tune -up.
The signal may now be monitored in a
nearby 50 Mc. receiver. Connect a dynamic
microphone and modulate the transmitter, adjusting the audio gain control for good modulation. The transmitter is now ready to be
connected to your station antenna.
28 -2
581
styling, this deluxe unit is designed around the
7270 beam power tube and is capable of a
conservative input of 200 watts on phone,
and 250 watts on c.w. The transmitter covers
all amateur bands between 10 and 80 meters,
is v.f.o. controlled, and incorporates speech
clipping for maximum audio "punch." A semiconductor high voltage rectifier is used to
reduce heat and to provide improved voltage
regulation. "Break -in" c.w. keying is incorporated employing a time differential system that
results in chirp-free, clickless keying. Band
changing is simplified by ganging the exciter
switching circuits with the final amplifier pinetwork so that single control adjustment is
achieved. In short, the transmitter incorporates
all modern techniques to make it an up -todate, valuable item of station equipment that
will not become obsolete.
Circuit
Description
A block diagram of the table top transmitter is shown in
figure 8. Thirteen tubes are
five in the r.f. section, five
employed,
in the audio section, and the remainder in the control and power supply section. A complete schematic is shown in
figures 9 and 10. The RCA 7270 beam power
tube is employed in the final amplifier stage.
This compact tube has high -perveance and
good power gain. It can be operated at full
input above 50 Mc., and has a maximum plate
dissipation of 90 watts. At a plate potential
of 1000 volts, this miniature "bottle" is capable of 250 watts input on c.w., and 200 watts
input on a.m. phone. In addition, the tube has
triple base -pin connections for the screen grid
to permit good r.f. grounding and has large
plate radiating fins for effective cooling. The
A Deluxe
200 -Watt Tabletop
Transmitter
This self contained, TVI- proof, tabletop
transmitter is designed for the amateur who
desires a compact station capable of running
sufficient power to provide consistent results
in today's busy amateur bands. Modern in
Figure 6.
VOLTAGE REGULATED POWER SUPPLY.
The silicon rectifiers are mounted above the
chassis for proper ventilation, with the two
transistors directly in front. 2N561 transistor
is insulated from the chassis by a mica shim.
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582
T H
Low Power Transmitters
compact size makes it especially effective in
the high frequency portions of the communication spectrum. Driving requirements are
modest and permit the use of a simple band switching exciter.
The Exciter Section. The high stability, all band v.f.o. consists of a 6AH6 (V1) in a
"hot cathode" circuit, followed by a 6CL6
(V2) crystal oscillator- buffer stage. Very
high -C is used in the oscillator stage to swamp
out variations and changes in stray circuit and
tube capacitance. The frequency determining
circuit operates on 80 meters (L1 and associated components) for 80, 40 and 10 meter
transmitter operation, and on 40 meters (L_
and associated components) for 20 and 15
meter transmitter operation. The circuit is a
modified version of the Clapp oscillator. The
tuning rate for each amateur band is changed
automatically so that each band is spread over
the entire portion of the tuning dial. Use of
the exceptionally smooth Eddystone dial with
a turn indicator makes it possible to read the
transmitter frequency within a kilocycle or
two. The oscillator is keyed by a section of the
12ÁU7 keyer tube (V6) for c.w. operation.
The crystal oscillator- buffer stage (6CL6,
V_) employs a broadly tuned 7 Mc. plate
circuit for operation on 40 meters and all
E
R
A D
I
O
higher bands. For 80 meter operation, switch
section SIB inserts an r.f. choke in series with
the tuned circuit, dropping the r.f. output on
this band to the correct value, and eliminating
the necessity of tracking the stage across the
relatively wide band. Switch S2 disables the
v.f.o. and converts tube Vo into a 3.5 Mc.
crystal oscillator, with the choice of two
crystal frequencies.
Figure 7.
MODERN 200 WATT
ALL BAND TABLE -TOP TRANSMITTER.
Complete TVI -proof phone and c.w. transmitter is housed in modern -style tabletop cabinet.
V.f.o. controlled, the transmitter covers all
amateur bands between 80 to 10 meters. High
level plate modulation with speech clipping is
used fcr ptione, and a time -sequence break -in
keyer is featured for c.w. operation. A standard 10 /s" x 19" panel is used in cose rock
mounting is desired. Multi -meter on the left
reads grid and screen current of amplifier
stage, or modulator plate current. Selector
switch is at left, directly below main tuning
dial.
Controls across bottom of panel are (I to r.):
audio -gain, microphone receptacle, filament on switch, amplifier plate tuning (top) and
amplifier plate loading (bottom), bandswitch,
v.f.o.- crystal switch (top) and amplifier arid
tuning (bottom), power switch (Ss) and pilot
light, c.w.- tune-a.m. switch, and key jack.
Below the tuning dial to the right is the grid
drive control, and at the for right is the
plate meter, M2.
www.americanradiohistory.com
HANDBOOK
VI
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200 -Watt Transmitter
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BLOCK DIAGRAM OF
Figure 8.
200 WATT TABLE -TOP TRANSMITTER.
The plate circuit of the 6CL6 multiplier
stage (V3) is untuned for 80 and 40 meter
operation, and is resonated to 14 Mc. for 20
and 10 meter operation by coil L4, and to 21
Mc. for 15 meter operation by coil L5. This
stage is block -grid keyed for c.w. operation.
A 2E26 (V4) is used as a driver for the
7270 amplifier. This stage is neutralized and
operates "straight through" on all bands except
10 meters, where it acts as a doubler from
14 Mc. A potentiometer control (grid drive)
in the screen circuit of the 2E26 determines
the excitation level to the final amplifier
stage.
The 7270 (V3) serves as a neutralized
amplifier on all bands. Grid, screen and plate
current are monitored for proper operation. A
pi- network output circuit permits operation
into unbalanced loads having impedances in
the range of 50 to 75 ohms, and an s.w.r. value
of 2.5 to 1, or less. The screen circuit is protected by relay RY3 which is energized by
application of primary power to the high voltage plate supply. Thus, screen voltage cannot
be applied to the tube unless plate voltage is
also applied.
The Mode Switch, S3. For tune -up purposes,
amplifier screen voltage is dropped to a low
value by the c.w.- tune -a.m. mode switch section S3,,. In the c.w. position, protective cut -off
bias is applied to the 7270 by switch section
S{i,. For phone operation, the amplifier screen
circuit is "self- modulated" by choke CHI
placed in the circuit by switch section Sac.
The keyer tube (V11, 12AU7) keys the
oscillator in addition to the 6CL6 multiplier
stage, and optimum break -in characteristics
may be set by the variable potentiometer
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THE RADIO
Low Power Transmitters
586
labelled adjust keying. Switch section S3A
shorts out the keyer in the tune and a.m.
modes. Switch sections StD and S3F disable
the modulator and speech amplifier in the
c.w. and tune positions. Switch section S3E
activates the 400 volt power supply in the
Figure 11.
TOP VIEW
OF
200 WATT TRANSMITTER.
Layout of above-chassis ports is shown in this
view. Amplifier compartment is center rear,
with buffer compartment between it and v.f.o.
box near panel. Main power transformer is to
the left of amplifier compartment, with modulation transformer and 811 -A tubes to the
right.
Adjacent to the horizontal oscillator tube (on
left of v.f.o. box) are the two 6CL6 amplifier
stages. The low- voltage power supply components are on the separate chassis at the
left. On the side of the main chassis is the
large cutout for air intake to the ventilating
fan. To the right of the v.f.o. box are located
the 12AU7 keyer tube and the OB2 voltage
regulator. The adjust "keying" and "clipping"
controls are between these tubes and the
shielded speech amplifier tubes on the far
edge of the chassis. The 6DE7 is next to the
modulator tubes.
tune positions, and the supply is
activated by push -to -talk relay RY1 when
switch S3 is set in the phone position. In
addition, the transmitter may also be activated
by the panel mounted power switch, S5 which
completes the relay control circuit.
c.w. and
The Modulator Section. A pair of zero bias
811A tubes (V11, V12) are used as class
B modulators, eliminating the need of bias
and screen power supplies which are costly
and expensive. A dual -purpose 6DE7 (V10)
serves as a speech amplifier and driver stage.
A 6AL5 double diode (V9) is a low level
audio peak clipper which serves to increase
the average level of modulation. This stage
is followed by a home-made low -pass filter
that restricts all audio frequencies above 3000
cycles. A 12AX7 dual triode (V8) provides
sufficient gain for proper transmitter operation from a low level crystal microphone. In
addition, a microphone push -to -talk circuit
may be used to energize d.c. relay RY1 by
means of a microphone control switch.
www.americanradiohistory.com
HANDBOOK
200 -Watt Transmitter
Power Supplies. A careful selection of power
supply components makes it possible to build
a transmitter of this capability in such a
small space. A cooling fan has been incorporated to insure that proper movement of air
is maintained, and components have been
selected for adequate safety margins and cool
operation. The chassis has several cut -out
openings on the sides and top for air circulation, and the chassis bottom plate of the audio
section is made of perforated aluminum.
The low voltage and bias supplies are conventional; however, the high voltage supply
makes use of a bridge circuit employing twelve
miniature silicon diode rectifiers. The center
tap of the transformer high voltage winding
is not used, and the 5 -volt winding is employed only to light a panel indicator lamp
when the high voltage is switched on. The
high voltage rectifier "stack" is protected from
accidental overloads by a 1/2- ampere fuse
placed in the B -plus lead to the filter system.
Three 40 pfd., 450 -volt electrolytic capacitors are placed in series to provide approximately 12 izfd. at a working voltage of 1350.
The entire transmitter, including power supplies, is built
upon a heavy aluminum chassis
measuring 13" x 17" x 3" in size. Shielded,
chassis-type construction is used, and no reliance is placed upon the cabinet for TVIreduction (figures 7 and 11) . The v.f.o. is
Transmitter
Construction
built as a separate unit in a 3" x 4" x 5"
aluminum box which is bolted to the main
chassis behind the geared dial.
The low voltage and bias supplies are built
on an aluminum chassis measuring 4" x 7" x
11/2", and may be seen in figure 11. The
5V4 -GA rectifier tube (V13) is mounted
"outboard" on a small L- shaped bracket beside
the power transformer (T2), fitting in nicely
between the supply chassis and the buffer
stage. The supply leads are brought through
grommeted holes in the main chassis to terminal strips placed on the side apron of the
chassis.
The 7270 amplifier stage is entirely enclosed in an aluminum box measuring 61/2
inches square and 614 inches high (figure
12) . The top and back of this enclosure are
fabricated from a single piece of perforated
aluminum. The other three sides of the box
are formed from an aluminum sheet, while
the main chassis serves as the bottom of the
enclosure.
The 2E26 buffer stage is mounted between
the v.f.o. enclosure and the final amplifier
compartment. The buffer tube is placed in a
horizontal position to best isolate the input
and output circuits, and to obtain short leads
in the plate circuit. The enclosure has screened
sides and measures 3" x 2" x 1/2 " in size.
The 2E26 tube projects into the box, with
the base connections remaining outside the
box in close proximity to the neutralizing
Figure 12.
CLOSEUP OF FINAL
The
AMPLIFIER
ASSEMBLY.
top and front of the
final amplifier enclosure
been removed to
show placement of major
components. The tank
coil is mounted in a ver-
have
tical position, bolted to
the side wall of the box.
The output loading capacitor is just below the
section.
capacitor is mounted on inbetween
pillars
sulated
the 7270 tube and the
tank tuning capacitor.
Plate leads are made of
10 -meter
The
coil
neutralizing
silver -plated
587
copper
strap. The perforated
shield at front of photo
covers the horizontally
2E26
buffer
mounted
tube.
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www.americanradiohistory.com
200 -Watt Transmitter
capacitor. The plate coils of the 2E26 are beneath the chassis, grouped about bandswitch
section S11). The buffer tuning capacitor
( labelled grid tuning) is adjacent to the bandswitch ( figure 13) .
Placement of the major components may be
seen in figure 11. The audio section is on
the right of the chassis (viewed from the
rear) and is separated from the r.f. section by
a partition running the entire depth of the
chassis on the underside. The 0A2 voltage
regulator tube (V7) and the 12AU7 differential keyer tube (V6) are also in the audio
section.
The remainder of the smaller components
are mounted beneath the chassis ( figure 13 ).
The modulator section is to the left, while
along the opposite side of the chassis are
located the small blower fan, the high voltage
silicon rectifiers, and the large filter choke.
The center portion of the chassis is reserved
for the r.f. section of the transmitter. The 7270
socket is centered towards the rear, directly
behind a horizontal partition that separates the
final amplifier components from the exciter
stages. Vent holes are cut in the side aprons
of the chassis (figure 11) and are covered
with screening.
In order to mount the 6.3 volt filament transformer (T4) on the side apron, a hole is drilled
in the side of the case and the transformer
leads are brought out through this hole, instead of via the bottom hole. This same technique is used to mount the high voltage filter
choke, CH_. To facilitate mounting these
components, 6 -32 nuts are soldered to the
mounting flanges to accept the mounting bolts.
The panel layout is dictared by placement of the
major components. The
v.f.o. tuning dial is centered on the panel near the top to allow proper
clearance for the drive mechanism. The dial
drive shaft, therefore, determines the position
of the dual v.f.o. tuning capacitor which is
mounted inside the enclosed oscillator assembly. A flexible coupling is used to join the
dial to the capacitor to provide proper shaft
alignment and smooth tuning. The v.f.o. itself
is built as a separate unit after the position
of the oscillator tuning capacitor has been
determined.
Panel Layout and
Bandswitch
Placement
589
The power amplifier output loading capacitor, plate tuning capacitor, and pi- network
coil switch (S1E) are controlled from the
front panel by means of right -angle drive systems placed beneath the chassis. The bandswitch S1 (centered on the panel) drives the
v.f.o. bandswitch through a right angle coupler,
in addition to driving the pi- network switch
of the amplifier stage. Two small bevel gears
are used for the oscillator drive, one mounted
on the main bandswitch assembly between
segments SIB and SID, and the other placed
on the shaft of switch SIA which is located in
the v.f.o. compartment (figure 14) . The oscillator bandswitch is placed directly below the
v.f.o. tuning capacitor, with its shaft on the same
vertical center line as that of the capacitor. The
switch projects down through a 3A-inch matching hole in the chassis, placing the shaft at
right angles to the center line of the main
bandswitch where it is driven by the bevel
gears.
The main bandswitch assembly passes along
the center line of the chassis to the final amplifier area, extending through a shield partition which isolates the multiplier and driver
coils. An added section of shaft coupling drives
a set of Boston gears mounted on a small support bracket at the back of the chassis. The
gears have a 1:2 step -down ratio, as the final
amplifier bandswitch has 60- degree indexing,
whereas the main bandswitch has 30- degree
indexing.
It is a good idea to assemble the chassis,
panel, and v.f.o. box, and lay out the various
gear drive systems before other holes are
drilled or components mounted in place. The
final amplifier tuning and loading capacitors
are mounted alongside the pi- network coil
and their shafts project into the under -chassis
area where they are joined to right -angle
drives which bring the controls to the front
panel. The amplifier tuning capacitor is driven
with a set of Boston gears having a 2:1 step up ratio so that the dial turns 360- degrees
while the capacitor rotates 180- degrees. This
makes for easier adjustment of the circuit.
Placement of the remaining panel controls
and meters is not critical and is dictated by
good symmetry and eye -appeal. Panel and
chassis should be drilled together so that all
shaft holes are in alignment. Panel and chassis
are held together by two 13 -inch aluminum
angle brackets placed at the ends of the chassis.
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590
Low Power Transmitters
THE RADIO
Figure 14.
INSIDE THE V.F.O. ENCLOSURE.
The oscillator tube socket is mounted to the left wall of the box, with the tuning capacitor
adjacent to the terminals. The two one-inch diameter ceramic coil forms are mounted to the
opposite wall with the padding capacitors between them. At ftc bottom of the box is the
oscillator bandswitch, driven from the main bondswitch below deck by right -angle gears. Extra
bolts are used to fasten the sides of the box securely in place, and all paint is scraped off the
mating areas to ensure good contact.
www.americanradiohistory.com
HANDBOOK
200 -Watt Transmitter
591
.0J.1 F
1
600V.
L
r--
EA.
7.001
47
K
1.0 JJF
600
l
V.
K7',4K
[CH 5
1
i100
47K
.01,600V.
1
K
.OJJF
600
Figure 15.
V.
time to obtain a spacing of about 3/16 inch, leaving nine plates in all (4 rotor, 5
stator). The capacitor is connected to the low
potential (pi- network) side of the plate blocking capacitor so that d.c. plate voltage does not
appear across it.
Oscillator Construction. The whole v.f.o. unit
may be wired separate from the transmitter.
The tuning capacitor is a dual 25 µpfd. unit,
with two rotor plates removed from the front
section which tunes the 40 meter coil (L0).
Two ceramic coil forms are mounted on the
wall of the v.f.o. box opposite the tuning
capacitor and two MAPC-type adjustable padding capacitors are in a line between the coils.
The oscillator tube socket is on the side wall
below the tuning capacitor, and all associated
resistors and capacitors are mounted at the
socket, with the exception of the silver mica
capacitors which make up the various tuned
circuits. These are mounted on the band switch, or to the wires running between switch
a
SPEECH AMPLIFIER TERMINAL BOARD.
Make of phenolic, or other insulating material.
The transmitter is most easily
worked upon if the heavy
and Wiring
transformers are left off the
chassis until the very last. The
v.f.o. and low voltage supplies can be wired
and tested as separate units before they are
affixed to the chassis.
The socket for the 7270 tube is recessed so
that the vent holes in the base are on the
underside of the chassis for passage of air
from the cooling fan. The variable neutralizing
capacitor for the amplifier stage is mounted
vertically between the socket and the plate
tuning capacitor (figure 12) and is adjustable from beneath the chassis. Space is limited
so a modified APC -type unit is used. It is
a 50 µpfd. size, with plates removed two at
Assembly
270
L_
J
Transmitter
'
www.americanradiohistory.com
amplifier components (including the audio
filter) are pre- assembled on a phenolic board
which is mounted on the side apron of the
chassis ( figure 15) . Small components are
soldered directly to socket pins. Miniature
transistor-type cathode bypass capacitors are
used to conserve space. The clipping and
keyer controls are mounted on the chassis deck,
between the low level stages. Filter inductor
(CH5) is made from a Stancor TA -27 audio
transformer, using the entire secondary winding as the coil. The voltage dividers of the bias
supply and the dropping resistor for the regulator tube are placed in this section.
The Power Supply and Control Circuits. The
5+
CUT OUT (WARD
2-}"A
2"
RECTIFIER BOARD
FRAMEWORK
TO N.V.
3
TO N.V.
ALUMINUM NEAT BARRIER
s
204 204 204
I.
THE RADIO
Low Power Transmitters
592
401r
450
V.
40Ur
4010
450V.
450 V.
I
6-12
FILTER BOARD
Figure 17.
LAYOUT OF RECTIFIER AND
FILTER BOARDS.
and coils. All tuned circuit wiring is done
with #14 solid tinned copper wire. The v.f.o.
output lead to the next stage passes via a feed through insulator in the bottom of the box to
the under -chassis area. Filament and power leads
are brought out through a grommet to a
terminal strip beneath the main chassis.
The Exciter and Audio Circuits. The exciter
wiring is straightforward. The slug-tuned exciter coils are grouped about the main band switch, and all r.f. leads are short and direct.
All r.f. bypass capacitors are mounted directly
on the socket terminals. Part of the speech
twelve silicon diode rectifiers and the filter
network are placed below the high voltage
transformer. The diodes are mounted on a
perforated frame made from a sheet of fiberglass or phenolic material (figure 17) . The
diodes are supported by their leads from small,
hollow rivets employed as connecting points.
The diode leads should be left untrimmed, and
the leads are grasped with a pliers during the
soldering process to prevent the heat of the
iron from injuring the diode. The diode
mounting plate is attached to the side apron
of the chassis in front of the large air vent,
directly behind the ventilating fan.
The main filter capacitor consists of three
series connected 450 volt capacitors in parallel
with three wirewound resistors. These components are wired as a unit and mounted on
a phenolic board on the rear apron of the
chassis, alongside the blower motor. The I/2ampere high voltage fuse holder is also
mounted on this board. A small aluminum
shield is placed between the resistors and
capacitors to act as a heat barrier ( figure 17)
Figure 16.
COIL DATA.
TABLE TOP TRANSMITTER.
L1-40
turns #22 enameled wire on
-17
turns #20 enameled wire on
I"
ceramic form
I"
ceramic form
(National XR -62) Range: 3.5 -4.0 Mc.
L2
(Notional XR -62) Range: 7.0 -7.175 Mc.
L3-40 turns »28 enameled wire
on
(National XR -50) Ronge: 7.0 Mc.
L4
-20
/2" form
i
form
turns #20 enameled wire on /2"
(National XR-50) Range:
14 Mc.
-12
turns »20 enameled wire on /2" form
(National XR -50) Range: 21 Mc.
L6 -16 turns 3/4" diameter tapped at 9 and 12
turns from junction with LT
( #3011
B&W miniductor)
L7-38 turns #24 tapped at center 1" diameter
( »3016 B&W miniductor)
LB
851 B&W tank coil assembly
L5
-»
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HANDBOOK
200 -Watt Transmitter
The a.c, line fuse holders, antenna connector,
relay connector J3, and Hy-pass feedthrough
capacitors for the power line are also mounted
on the rear apron. The coaxial antenna relay
(RY2) is placed on the outside of the apron
with a right-angle fitting added so that the
antenna connection is accessible when the
transmitter is placed in its cabinet.
Most of the power and control wiring follows along the front inside edge of the chassis.
Shielded wire is used for the 7270 filament
and screen leads, and filament circuits are
wired with #14 wire. The screen capacitor of
the 7270 stage consists of three separate .001
pfd., 3 kv. ceramic disc capacitors, one placed
from each screen socket pin to ground.
The multi -meter (M1) has a 5 milliampere
movement, and is converted into a low range
voltmeter by the addition of a 750 -ohm series
resistor. The voltage drops across shunts placed
in the grid and screen circuits of the amplifier, and the plate circuit of the class B modulator are measured in this fashion. The meter
scale is 0 -10 milliamperes when switch S4 is
in the grid position, 0 -40 milliamperes in the
screen position, and 0 -400 milliamperes in the
modulator position.
Tuning and
Adjusting the
Transmitter
When the transmitter is corn pleted, the wiring should be
visually inspected, and circuits
"rung out" with an ohmmeter. The next step is to test and adjust the
v.f.o. The fuse should be left out of the primary
circuit of the high voltage supply to disable
this section and to ensure that relay RY3 remains open. The 2E26 screen control should
be set to remove screen voltage. Starting with
the 80 meter band and the v.f.o. dial set at
the low end (maximum v.f.o. tuning capacitance), trimming capacitor C4 is adjusted for
3500 kc., as noted on a frequency meter. With
the specified coils, the 80 meter band extends
over the entire dial scale, with the slug almost
out of the coil form. Once the coverage is set,
the slug should be secured with an extra nut
to prevent movement. Capacitor C4 should not
be moved now, as it will be in the circuit for
the 40 and 10 meter adjustments. Next, the
bandswitch is placed in the 10 meter position,
the dial set at the low frequency end, and
trimmer capacitor C7 adjusted for 28.0 Mc.,
with the v.f.o. dial pointer in approximately
the same position as for 3.5 Mc. The 10 meter
593
band will now extend over almost the entire
scale. Next, the bandswitch is placed in the
40 meter position, and it will be noted that
the 7.0 Mc. position will fall very near the
28 Mc. mark, with the 40 meter band spread
over most of the scale.
The bandswitch should now be placed on
the 20 meter position, and trimmer capacitor
Cs adjusted so that 14.0 Mc. falls near the
3.5 Mc. dial point. The 15 meter calibration
is automatically set by this adjustment.
Once the v.f.o. has been calibrated, the 80
meter exciter circuits are tuned by simply advancing the 2E26 screen voltage potentiometer
and tuning the driver stage to resonance, as
indicated by grid current of the 7270 tube.
Grid current should be held to a maximum of
4 milliamperes. The bandswitch may now be
set to 40 meters and buffer coil L3 adjusted
for maximum amplifier grid current with the
v.f.o. set at 7.15 Mc. The 14 Mc. adjustments
are now made with the bandswitch in the 20
meter position, and coil L4 peaked for maximum amplifier grid current at 14.15 Mc.
Finally, the bandswitch is set to 15 meters and
the slug of coil L, is peaked at 21.2 Mc. The
driver stage is, of course, resonated for each
band. The ten meter band is tuned by merely
peaking the driver stage. Check both the low
and high ends of the 10 meter band and
equalize the grid drive by slight adjustments
to coils L3 and L4,
The 2E26 stage is neutralized in the 20
meter position by placing a temporary grid
meter in series with the "cold" end of the 22K
grid resistor and adjusting the neutralizing
capacitor for minimum meter kick when the
plate circuit is tuned through resonance.
Screen voltage should be removed for this test.
This setting will hold for all bands. Grid
current to the 2E26 should not run over 3
milliamperes.
The amplifier stage is now neutralized in a
similar manner, using meter M1 to observe
action of the grid current. This adjustment
should be done on the 10 meter band.
The final amplifier should not be operated
without a dummy antenna load of some kind.
Two 100 -watt lamp bulbs in parallel at the
end of a short length of coaxial line will make
a satisfactory load for preliminary adjustment
purposes. Place the high voltage primary fuse
in its receptacle and set the function switch S3
to the tune position. Grid current will now be
www.americanradiohistory.com
THE RADIO
Low Power Transmitters
594
NOTE: DIMENS IONS OF FLANGE TO FIT
TUN /NG CAPACITOR TERMINALS.
CAPACITORS BU /LT WITHIN
E /MAC SOCKET.
C
-
s
I
L
I
ANT.
A
+
SCR.
"PLATE" LINE Lx
5001
I
IBIAS ,00
DRILL BOTH PLATES FOR INSULATED
BOLTS AND BUSHING.
D
E %C.
L SUB -CHA SS /S
AREA.
F
ROUND
CORNERS
B+
STRIP LINE CAVITY
LINE WITH
FINGER STOCK
',CHOKE" LINE LI
ANT
DIMENSIONS
A
10
,Et/i11L'tB
ByyCyyDyy
The
E
Cy/D
BMCI
2i
220 MC.
144 MC
F
7}
A
E
21"
RFC,
F
4 -
EQUIVALENT CIRCUIT
Figure 18.
SCHEMATIC AND EQUIVALENT CIRCUIT OF STRIP LINE AMPLIFIER.
strip line amplifier is built within 3" x 5" x 13" aluminum chassis box (144 Mc.), or
2" x S" x 91/2" (220 Mc.). The plate tuning capacitor of the 144 Mc. assembly is a cut -down
turn,
Johnson 154 -11 having three plates, spaced 0.25" apart. The antenna "hairpin" loop is
4" long and 11/2" wide (144 Mc.), or 2" long 34" wide (220 Mc.) placed parallel to strip line.
Antenna resonating capacitor C2 is 35 µold. for either amplifier. Plate choke RFC, is Ohmite
1
or Z -220. B -plus lead posses through insulated hole in chassis, or may pass through
feed- through type capacitor for low voltage operation (500 volts or less). The screen bypass
capacitors are built within the Eimac air system sockets. Input circuits and blower are placed in
sub- chassis enclosure.
Z -144
observed on the final stage. The power switch
S5 is turned on energizing relay RY1, and the
final amplifier resonated and loaded to a plate
current of about 150 milliamperes. The series
screen resistor used in the tune mode limits
off - resonance amplifier plate current to less
than 200 milliamperes. The screen voltage tap
on the 2500 ohm, 25 watt resistor is now adjusted (with the transmitter off!) to place
about 320 volts on the 7270 screen circuit
with the function switch in the a.m. position,
and the amplifier loaded to 200 milliamperes
plate current. In the c.u'. position the screen
voltage will be slightly higher.
Maximum voltage (400 volts) is always
applied to the plate of the 2E26, and the dropping resistors reduce this to about 260 volts
for the v.f.o., 6CL6 stages, and speech amplifier. The final plate voltage runs 1000
volts at a load current of 200 ma., and rises
to about 1200 volts in the c.w., key -up position. Oscillator screen voltage is regulated at
105 volts. The bias supply delivers -135
volts, and the push -to -talk relay circuit is
tapped down on the bleed resistor to supply
about 100 volts to the d.c. relay RY1.
The c.w. keying characteristic is determined
by the adjustment of the keyer potentiometer,
and by the choice of the 0.1 pfd. capacitors in
the grid returns of the keyed tubes. For break in keying the "key -up" signal is monitored in
the receiver and the keyer potentiometer is
backed off until the oscillator signal just
disappears.
For phone operation, the modulator resting
plate current is about 20 ma., kicking up to
approximately 175 ma. on voice peaks. Maximum current excursions and modulation level
are set by the adjust clip control, and the
degree of modulation by the audio control.
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HANDBOOK
28 -3
Strip Line Circuit
Strip -Line
Amplifiers for VHF
Circuits
A major stumbling block in the design and
construction of high power v.h.f. transmitting
equipment is the assembly of a suitable amplifier plate tank circuit. Simple L -C tuned
circuits tend to assume microscopic proportions in this region of the spectrum and are
incapable of handling large amounts of r.f.
energy. Coaxial circuits, on the other hand,
work well but are expensive, difficult to
build and bulky to handle.
A welcome compromise design is the simple
strip line tank circuit, illustrated in figure
18A. The circuit is a modified cavity, making
use of an inexpensive aluminum chassis as
the outer enclosure, and employing strips of
aluminum as the plate inductance. The line
assumes r.f. ground potential at the end opposite the tube and is an approximate electrical
eighth -wavelength long. It becomes an electrical quarter -wavelength when loaded by the
tube and tuning capacitor placed at the high
impedance end of the line. The line is made
of two aluminum plates, separated by insulating material. This "sandwich" may be visualized as the equivalent circuit of figure 18B,
which permits plate voltage to be applied to
the amplifier tube via "plate line" L1 yet isolates the tuning capacitor and plate inductance
from the d.c. voltage by means of a "distributed" r.f. choke. The cavity is completed by
placing an aluminum cover plate over the
open side of the chassis.
A proper ratio of strip length and width
compared to the cavity dimensions must be
observed to determine the optimum line impedance, but the parameters may be varied
sufficiently to permit the use of an inexpensive ready -made chassis for the line cavity
without appreciable circuit degradation. Efficiency of the strip line is high, comparing
favorably with conventional tank circuits
operating at intermediate frequencies.
The approximate characteristic impedance
of the strip line may be determined from the
following formula:
377 S
(1)
Z
"="
W
where S is the spacing between the strip line
and the chassis, and W is the width of the
strip; the width being much greater than the
595
spacing. A practical strip line will be shorter
than a quarter wavelength by virtue of the
interelectrode capacitance loading of the associated tube and the auxiliary tuning capacitor
placed across the line (figure 18A) . In this
case, the characteristic impedance of a loaded
strip line is approximately:
(2)
Z = Z0 tan 131
0
p =
, ) is the wavelength
in
centimeters and 1 is the length of the line in
centimeters.
If the total capacitive reactance is set equal
to Z0, then tan /3, = 1, when the line length
is 1/2- wavelength. For example: Assume a
1/2- wavelength line having a width (W) of
3 inches and a spacing (S) of 1 inch. The
impedance, Zo is therefore (by formula 1)
about 127 ohms. The output capacitance of
a 4X250B is approximately 5 µµfd., representing an impedance value of about 220 ohms at
144 Mc. A parallel tuning capacitance of 5
µµfd. has the same impedance value, and the
combined parallel impedance is approximately
110 ohms. Therefore a 1/2- wavelength line of
the aforementioned dimensions could be used
to tune the 2 meter band with a 4X250B
tube. This line would be about 10 inches long,
so a standard chassis box measuring 3" x 5"
x 13" could be used for the plate cavity
assembly. The construction of such a unit is
described in this section.
where
Building the
Strip Line
3
Shown in figure 19 are two
strip line units for 144 Mc.
Circuit
and 220 Mc. The amplifiers
are designed around the ceramic 4CX250B tube and may be operated at
power inputs up to 500 watts for c.w. service,
or 300 watts for a.m. phone. The limiting
factors for power input are the maximum
voltage rating of the plate bypass capacitor
(if used), tuning capacitor spacing, and the
voltage breakdown of the material employed
as the dielectric of the strip line circuit.
The units illustrated employ 10 -mil teflon
coated fiberglass as the strip line dielectric,
with fiber or teflon bushings and 4 -40 machine
screws holding the assembly together. It is
also possible to purchase teflon screws which
could be used to advantage in this assembly.
A sheet of 10 -mil mylar may be substituted for
the fiberglass.
Layout of the strip line units is shown in
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596
THE RADIO
Low Power Transmitters
Figure 19.
STRIP LINE AMPLIFIERS FOR 144 MC. AND 220 MC.
The simple mechanical assembly of the strip line tank circuit is especially suitable for home construction. Using o standard aluminum chassis as the foundation, the strip line consists of two
aluminum plates separated by a dielectric. The line is supported from one end of the chassis
box, and the tube socket is mounted in the bottom, with the tuning capacitor at the opposite
end. At the near end of the assembly are the antenna resonating capacitor, the B -plus terminal
and the antenna coaxial receptacle. The tube plate "finger stock" connector is made by EitelMcCullough, Inc., San Carlos, California, part =008294, Anode Collet.
The dielectric material for the "sandwich" may be either 10 -mil (0.01 ") Mylar sheet, or 10 -mll
teflon coated fiberglass. The mylar may be obtained from: Milam Co., 1100 Elmwood St., Providence, R. I. The teflon coated fiberglass may be obtained from Dodge Fibers, Inc., Hoosick Falls,
N. Y. For maximum values of plate voltages, two layers of material should be used. Open side of
chassis is closed by cover plate.
figure 18. The "plate" section of the line (L2)
is bolted to one end of the chassis box, at the
proper height to encircle the anode of the tube
without actually touching it. The "hot" end
of this line is affixed to the stator of the plate
tuning capacitor. The capacitor of the 144 Mc.
amplifier has 0.25" spacing, as the unit is designed for high power operation. The "choke"
plate of the "sandwich" line (L1) is shorter in
length and spaced away from the grounded
plate by means of the sheet fiberglass or mylar
insulator. One end of this plate is connected
to the B- supply through an auxiliary r.f. choke,
and the opposite end makes contact to the
anode of the tube by means of flexible metal
finger stock soldered to the plate (see parts
list) Both plates are sanded smooth to ensure
that no metallic splinters or grains can puncture the thin dielectric sheet.
The 220 Mc. unit is designed for low power
doubler service at 500 volts and therefore
makes use of a receiving -type capacitor in the
plate circuit. A capacitor having greater spacing would be required for high voltage
operation.
The strip line amplifiers employ standard
Eimac v.h.f. air sockets to ensure stability of
operation. A standard grid circuit is employed
and if neutralization is desired, it is possible
to insert a probe into the strip line cavity and
.
www.americanradiohistory.com
HANDBOOK
"9T0" Electronic
feed back a small amount of energy in the
proper phase to the grid circuit. A "hairpin"
loop (L3) provides coupling to the antenna
circuit, and the reactance of the loop is tuned
out by means of a series capacitor. The grid
circuit components are built within a small
chassis box placed beneath the strip line assembly, with a cooling blower mounted on
the side of the box.
The dimensions given are correct for the
4X150A- 4CX250B type tube, but may be
varied for other tubes having slightly different interelectrode capacitances. Length of the
strip line and the value of the tuning capacitor
determine the resonant frequency, with the
width of the center line and chassis spacing
determining line impedance and exhibiting a
second order frequency effect. It is therefore
possible to effect small changes in the frequency of the circuit by varying the value of
the tuning capacitor or the width and chassis
spacing of the line if it is mechanically
awkward to adjust the length of the strip.
28-4
A "9T0"
Electronic Key
The good c.w. operator is always trying to
improve his skill and increase his keying
speed. The modern way to do this is to use
an electronic key. The dots, dashes, and spaces
are all created electronically with a minimum
Key
597
of effort on the part of the operator. A good
keyer has a "mechanical mind of its own" and
almost teaches the operator to send good code!
Shown in this section is a version of the
famous "9T0" keyer which provides the ultimate in reliable, precise electronic code. The
keyer uses four tubes and two voltage regulators, and is packaged in a cabinet only
slightly larger than a mechanical "bug" key.
Best of all, it is inexpensive to build and
fool -proof in operation.
Operation
of the Keyer
One of the most reliable and
stable methods of generating
automatic and self- completing
dots and dashes is the multivibrator system
used in this keyer (figure 21). The keyer is
driven by a "sideswiper" key which completes
a control circuit to ground in either the "dot"
or "dash" position. Closing the key on the
"dot" side energizes the dot keyer tube
(VOA) which turns on the dot multivibrator
tube (VIA_B) to form a string of evenly
spaced dots. Once the action has started, this
generator will continue to form dots as long
as the key contact is closed and will complete
a full dot even if the key is released in the
middle of a dot or a space. The output of the
dot multivibrator is fed to the grid of the
relay tube (VIA), and the contacts of the
quick- acting relay in the plate circuit are used
Figure 20.
THE "9T0"
ELECTRONIC KEY.
simple, inexpensive
electronic key generates
dots, dashes, and spaces
with a minimum effort
on the part of the operator. Four tubes and two
voltage regulators are
used in a simple and reThis
liable circuit. The "side swiper" key is mounted
to an extension of the
bottom plate of the
keyer, making a unit
only slightly larger than
"bug" key.
Panel controls are (I. to
a mechanical
r.): Weight control (with
on -off switch), monitor
speaker and speed control. Below these are the
or
zero -beat,
tune-up
button, and the neon
character indicator.
www.americanradiohistory.com
THE RADIO
Low Power Transmitters
598
+150
22
0.
VIA
-12AU7
1,200
V.
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MATCHED PAIR RESISTORS
ALL RESISTORS
V2A
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2200
21e
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OTHERWISE NOTED
M+
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108V
SR2
Figure 21.
SCHEMATIC, ELECTRONIC KEY.
RYI -DPST, 5000 ohm relay. Potter -Brumfield SM -SLS. Other satisfactory (but larger)
relays are: Claire HG-1002 or W. E. 2766. The 15K series resistor may have to be
adjusted for different relay models. Weight of dots may be varied by changing
value of this resistor.
3-
SRI,
Silicon rectifier. p.i.v. 400 volts @ S00 ma. Sackes-Tarzian «M -500.
TI -150 v. @ 50 ma., 6.3 v. @ 2a. Stancor PA -8421
T2- Push -pull replacement output transformer. Stancor A -3856
Key- Autronic sideswipes. Electrophysics Corp., 2500 West Coast Highway,
Beach, California
to key the transmitter and to activate an audio
tone oscillator (V4B) used as a monitor.
When the key is closed in the "dash" position, the dash keyer tube (V.,B) is energized,
placing the dash multivibrator tube (V ;i A.B )
in readiness for operation, and at the same
time sending a pulse through the 1N34 diode
to start the dot multivibrator circuit again.
This, in turn, triggers the dash multivibrator,
Newport
turning it on with the start of the first dot
pulse, and turning it off with the end of the
second dot pulse. The dash multivibrator,
therefore, is an electronic switch which is
turned on and off by two dot pulses. A dash
of proper length and timing is created in this
manner because the time length of the second
"dot" adds to the "on" time of the switch circuit in holding the relay closed for the dash.
www.americanradiohistory.com
HANDBOOK
"9T0"
Electronic Key
599
Keyer Construction
and Wiring
Figure 23.
TERMINAL BOARD LAYOUT.
The parts shown outside of board are mounted
underneath it. The lines indicate connections
made to the board from tube pins or other
components.
The complete keyer configuration makes use
of four 12ÁU7 double triode tubes. The power
supply uses two silicon diodes to furnish both
a positive and a negative voltage, regulated by
the 0A2 and OB2 regulator tubes. Unregulated
voltage is supplied to the relay tube and the
tone oscillator. If desired, the 5963 computer type tube may be substituted for Vt, V., and
V3 for improved long term stability of
operation.
The electronic keyer is
built upon an aluminum
chassis measuring 6" x
4" x 2 ", having two
auxiliary end plates 5 inches high. A wrapover perforated aluminum cover screens the
top and sides providing maximum ventilation.
In addition, four large holes are punched in
the sides of the chassis for additional cooling.
The "sideswiper" key is mounted on an extension of the bottom plate of the chassis. The
wiring of the keyer is simplified by mounting
most of the multivibrator components on a
terminal board placed in the underchassis area
( figure 22)
The board is mounted on two
pillars in the front -center of the chassis after
all other wiring has been done (figure 23).
The balance control is mounted on the rear
apron of the chassis, as it requires adjustment
only at intervals as the tubes age.
When the unit is completed, all wiring
should be checked. The unit is turned on and
after a short warm -up period the key lever
is held in the dash position and the balance
control adjusted until self -completing dashes
are formed. The neon lamp will flash at the
character rate. The speed and weight controls
are adjusted to suit the individual taste of the
operator.
Figure 22.
UN DE RCHASSIS
VIEW OF KEYER.
The resistors and capacitors making up the mul-
tivibrators are mounted
on a terminal board sup-
+os,
004r 04.
ported below the chassis
on short pillars. The silicon diode power rectifiers are on the side
apron of the chassis adjacent to the filter capacitors.
The
balance
potentiometer is on the
rear apron between the
keying lead and the
.F')
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www.americanradiohistory.com
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Low Power Transmitters
Figure 24.
TOP VIEW OF KEYER.
The keyer is built upon a 4" x 6" aluminum chassis. Layout of parts is not crowded. The audio
oscillator transformer is near the front panel below the controls, and the sealed high -speed relay
is in the center of the chassis with the 12AU7 tubes on either side. The two regulator tubes are
between the power transformer and the rear panel.
www.americanradiohistory.com
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www.americanradiohistory.com
CHAPTER TWENTY -NINE
The trend in design of transmitters for operation on the high frequency bands is toward
the use of a single high -level stage. The most
common and most flexible arrangement includes a compact bandswitching exciter unit,
with 15 to 100 watts output on all the high frequency bands, followed by a single power
amplifier stage. In many cases the exciter unit
is placed upon the operating table, with a coaxial cable feeding the drive to the power amplifier, although some operators prefer to have
the exciter unit included in the main transmitter housing.
This trend is a natural outgrowth of the increasing importance of v -f -o operation on the
amateur bands. It is not practical to make a
quick change in the operating frequency of a
transmitter when a whole succession of stages
must be returned to resonance following the
frequency change. Another significant factor in
implementing the trend has been the wide acceptance of commercially produced 75 and
150 -watt transmitters. These units provide r-f
excitation and audio driving power for high level amplifiers running up to the 1000 -watt
power limit. The amplifiers shown in this
chapter may be easily driven by such exciters.
29 -1
Power Amplifier
Design
Choice of
Tubes
Either tetrode or triode tubes may
be used in high -frequency power
amplifiers. The choice is usually
dependent upon the amount of driving power
that is available for the power amplifier. If
a transmitter -exciter of 100 -watt power capability is at hand (such as the Heath TX-1)
it would be wise to employ a power amplifier whose grid driving requirements fall in
the same range as the output power of the
exciter. Triode tubes running 1- kilowatt input (plate modulated) generally require some
50 to 80 watts of grid driving power. Such
a requirement is easily met by the output
level of the 100 -watt transmitter which should
602
www.americanradiohistory.com
Design
ANTENNA
CIRCUIT
TO ANTENNA
TO
CIRCUIT
i
t
BALANCED TWIN- LINE
,FED SYSTEM
UNBALANCED COAXIAL
FEED SYSTEM
Figure
1
LINK COUPLED OUTPUT CIRCUITS
FOR
603
PUSH -PULL AMPLIFIERS
be employed as the exciter. Tetrode tubes
(such as the 4 -250A) require only 10 to
15 watts of actual drive from the exciter for
proper operation of the amplifier stage at 1kilowatt input. This means that the output
from the 100 -watt transmitter has to be cut
down to the 15 watt driving level. This is a
nuisance, as it requires the addition of swamping resistors to the output circuit of the transmitter-exciter. The triode tubes, therefore,
would lend themselves to a much more convenient driving arrangement than would the
tetrode tubes, simply because their grid drive
requirements fall within the power output
range of the exciter unit.
On the other hand, if the transmitter- exciter
output level is of the order of 15 - 40 watts
(the Johnson Ranger, for example) sufficient
drive for triode tubes running 1- kilowatt input
would be lacking. Tetrode tubes requiring low
grid driving power would have to be employed
in a high -level stage, or smaller triode tubes requiring modest grid drive and running 250
watts or so would have to be used.
Power Amplifier
Either push -pull or single
ended circuits may be emof Circuits
ployed in the power amplifier. Using modern tubes
and properly designed circuits, either type is
capable of high efficiency operation and low
harmonic output. Push -pull circuits, whether
using triode or tetrode tubes usually employ
link coupling between the amplifier stage and
the feed line running to the antenna or the antenna tuner.
It is possible to use the link circuit in either
an unbalanced or balanced configuration, as
shown in figure 1, using unbalanced coaxial
line, or balanced twin -line.
Design- C,oice
Figure 2
CONVENTIONAL PUSH -PULL
AMPLIFIER CIRCUIT
The mechanical layout should be symmetrical
and the output coupling provision must be
evenly balanced with respect to the plate coil
C:- Approx. 1.5 1I.Id. per meter of wavelength per section
Cr -Refer to plate tank capacitor design in
Chapter 11
Cr-May be S00 111.ld., 10,000 -volt type ceramic capacitor
NC -Max. usable capacitance should be greater, and min. capacitance less than rated
grid -plate capacity of tubes in amplifier.
50,-, greater air gap than C,.
R,-100 ohms, 20 watts. This resistor serves
as low Q r -f choke.
RFC,
-band r -f choke suitable for plate
current of tubes
M,
Suitable meters for d -c grid and plate
currents
All low voltage .001 pfd. and .01 pfd. by -pass
capacitors are ceramic disc units (Centrolab DD or equiv.)
L, -50 -watt plug -in coil, center link
L-.- Plug -in coil, center link, of suitable power
-All
-M-
rating.
Common technique is to employ plug -in
plate coils with the push -pull amplifier stage.
This necessitates some kind of opening for coil
changing purposes in the "electrically tight"
enclosure surrounding the amplifier stage. Care
must be used in the design and construction
of the door for this opening or leakage of harmonics through the opening will result, with
the attendant TVI problems.
Single ended amplifiers may also employ
link -coupled output devices, although the trend
is to use pi- network circuits in conjunction
with single ended tetrode stages. A tapped or
otherwise variable tank coil may be used which
is adjustable from the front panel, eliminating
the necessity of plug -in coils and openings
into the shielded enclosure of the amplifier.
Pi- network circuits are becoming increasingly
popular as coaxial feed systems are coming
into use to couple the output circuits of transmitters directly to the antenna.
www.americanradiohistory.com
604
H.F. Power Amplifiers
29 -2
Push -Pull Triode
Amplifiers
Figure 2 shows a basic push -pull triode
amplifier circuit. While variations in the method of applying plate and filament voltages and
bias are sometimes found, the basic circuit
remains the same in all amplifiers.
The amplifier filament transformer should be placed
right on the amplifier chassis in close proximity
to the tubes. Short filament leads are necessary
to prevent excessive voltage drop in the connecting leads, and also to prevent r -f pickup
in the filament circuit. Long filament leads
can often induce instability in an otherwise
stable amplifier circuit, especially if the leads
Filament Supply
are exposed to the radiated field of the plate
circuit of the amplifier stage. The filament
voltage should be the correct value specified
by the tube manufacturer when measured at the
tube sockets. A filament transformer having a
tapped primary often will be found useful in
adjusting the filament voltage. When there is
a choice of having the filament voltage slightly higher or slightly lower than normal, the
higher voltage is preferable. If the amplifier is
to be overloaded, a filament voltage slightly higher than the rated value will give greater
tube life.
Filament bypass capacitors should be low internal inductance units of approximately .01
p,fd. A separate capacitor should be used for
each socket terminal. Lower values of capacitance should be avoided to prevent spurious
resonances in the internal filament structure of
the tube. Use heavy, shielded filament leads for
low voltage drop and maximum circuit isolation.
The series plate voltage feed
shown in figure 2 is the most
satisfactory method for push -pull stages. This
method of feed puts high voltage on the plate
tank coil, but since the r -f voltage on the coil
is in itself sufficient reason for protecting the
coil from accidental bodily contact, no additional protective arrangements are made necesPlate Feed
sary by the use of series feed.
The insulation in the plate supply circuit
should be adequate for the voltages encountered. In general, the insulation should be
rated to withstand at least four times the maximum d-c plate voltage. For safety, the plate
meter should be placed in the cathode return
lead, since there is danger of voltage break-
down between a metal panel and the meter
T H
E
R
A
D
I
O
movement at plate voltages much higher than
one thousand.
The recommended method of obtaining bias for c -w or plate modulated telephony is to use just sufficient fixed
bias to protect the tubes in the event of excitation failure, and to obtain the rest by the
voltage drop caused by flow of rectified grid
current through a grid resistor. If desired, the
bias supply may be omitted for telephony if an
overload relay is incorporated in the plate circuit of the amplifier, the relay being adjusted
to trip immediately when excitation is reGrid Bias
moved from the stage.
The grid resistor R. serves effectively as
an r -f choke in the grid circuit because the impressed r -f voltage is low, and the Q of the
resistor is poor. No r -f choke need be used in
the grid bias return lead of the amplifier, other
than those necessary for harmonic suppression.
The bias supply may be built upon the amplifier chassis if care is taken to prevent r -f
from finding its way into the supply. Ample
shielding and lead filtering must be employed
for sufficient isolation.
The Grid Circuit
As the power in the grid
circuit is much lower than
in the plate circuit, it is customary to use a
close- spaced split-stator grid capacitor with
sufficient capacitance for operation on the
lowest frequency band. A physically small capacitor has a greater ratio of maximum to
minimum capacitance, and it is possible to obtain a unit that will be satisfactory on all bands
from 10 to 80 meters without the need for auxiliary padding capacitors. The rotor of the grid
capacitor is grounded, simplifying mounting of
the capacitor and providing circuit balance and
electrical symmetry. Grounding the rotor also
helps to retard v -h-f parasitics by by- passing
them to ground in the grid circuit. The L/C
ratio in the grid circuit should be fairly low,
and care should be taken that circuit resonance is not reached with the grid capacitor at
minimum capacitance. That is a direct invitation for instability and parasitic oscillations
in the stage. The grid coil may be wound of
no. 14 wire for driving powers of up to 100
watts. To restrict the field and thus aid in
neutralizing, the grid coil should be physically no larger than absolutely necessary.
Circuit Layout
The most important consideration in constructing a
push -pull amplifier is to maintain electrical
symmetry on both sides of the balanced cir-
www.americanradiohistory.com
HANDBOOK
Design
605
cuit. Of utmost importance in maintaining electrical balance is the control of stray capacitance between each side of the circuit and
ground.
Large masses of metal placed near one side
of the grid or plate circuits can cause serious
unbalance, especially at the higher frequencies, where the tank capacitance between one
side of the tuned circuit and ground is often
quite small in itself. Capacitive unbalance
most often occurs when a plate or grid coil is
located with one of its ends close to a metal
panel. The solution to this difficulty is to
mount the coil parallel to the panel to make
the capacitance to ground equal from each end
of the coil, or to place a grounded piece of
metal opposite the "free" end of the coil to
accomplish a capacity balance.
Whenever possible, the grid and plate coils
should be mounted at right angles to each
other, and should be separated far enough
apart to reduce coupling between them to a
minimum. Coupling between the grid and plate
coils will tend to make neutralization frequency
sensitive, and it will be necessary to readjust
the neutralizing capacitors of the stage when
changing bands.
All r -f leads should be made as short and
direct as possible. The leads from the tube
grids or plates should be connected directly
to their respective tank capacitors, and the
leads between the tank capacitors and coils
should be as heavy as the wire that is used
in the coils themselves. Plate and grid leads
to the tubes may be made of flexible tinned
braid or flat copper strip. Neutralizing leads
should run directly to the tube grids and plates
and should be separate from the grid and plate
leads to the tank circuits. Having a portion of
the plate or grid connections to their tank circuits serve as part of a neutralizing lead can
often result in amplifier instability at certain
operating frequencies.
Excitation
In general it may be stated
Requirements
that the overall power requirement for grid circuit excitation
to a push -pull triode amplifier is approximately
10 per cent of the amount of the power output
of the stage. Tetrodes require about 1 per cent
to 3 percent excitation, referred to the power
output of the stage. Excessive excitation to
pentodes or tetrodes will often result in, reduced power output and efficiency.
Push -Pull
Symmetry is the secret of suc Amplifier
cessful amplifier design. Shown
Construction
in figure 3 is the top view of
a 350 watt push -pull all band
LAYOUT
Figure 3
350 -WATT PUSH -PULL
TRIODE AMPLIFIER
OF
Two 811 -A tubes are employed in this circuit.
Plate tuning capacitor is at left of chassis,
with swinging -link type plug -in coil assembly
mounted above it. Rotor of split -stator capacitor may be insulated from ground to increase
voltage breakdown rating of capacitor. Note
that pickup link is series -tuned to reduce circuit reactance. One corner of rotor plate of
series capacitor is bent so that capacitor
shorts itself out at maximum capacitance.
Grid circuit coil and capacitor are at right.
Center- linked plug -in coil is employed. Parasitic chokes are placed in grid leads adjacent
to the tube sockets, and tube filaments are
bypassed to ground with .01 ,.fd. ceramic
capacitors. Complete area above the chassis
is enclosed with perforated screen to reduce
radiation of r.f, energy.
amplifier employing 811 -A tubes. The circuit
corresponds to that shown in figure 2 except
that the 811 -A's are zero bias tubes. The bias
terminals of the circuit are therefore jumpered
together and no external bias supply is required at plate potentials less than 1300 volts.
All r -f components are mounted above
deck. The plate circuit tuning capacitor and
swinging link tank coil are to the left, with
the two disc -type neutralizing capacitors between the tank circuit and the tubes. At the
right of the chassis is the grid tank circuit.
Small parasitic chokes may be seen between
the tube sockets and the grid circuit. Plate
and grid meters are placed in the under-chassis
area where they are shielded from the r -f field
of the amplifier.
Larger triode tubes such as the 810 and
8000 make excellent r -f amplifiers at the kilowatt level, but care must be taken in amplifier layout as the inter -electrode capacitance of
these tubes is quite high. One rube and one
neutralizing capacitor is placed on each side
of the tank circuit (figures 4 and 5) to permit
very short interconnecting leads. The relative
position of the tubes and capacitors is trans-
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606
Figure 4
UNIQUE CHASSIS LAYOUT PERMITS
SHORT LEADS IN KILOWATT
AMPLIFIER
Large size components required for high level
amplifier often complicate amplifier layout.
In this design, the plate tank capacitor sits
astride small chassis running lengthwise on
main chassis. Inductor is mounted to phenolic
plate atop capacitor. Variable link is panel
driven through right -angle gear drive. Plate
circuit is grounded by safety arm when panel
door is opened. Note that plate capacitor is
mounted on four TV -type capacitors which
serve to bypass unit, and also act as supports.
A small parasitic choke is visible next to the
grid terminal of the 810 tube.
posed on each side of the chassis, as shown in
the illustrations. The plate tank coil is mounted parallel to the front panel of the amplifier
on a phenolic plate supported by the tuning
capacitor which sits atop a small chassis -type
box. The grid circuit tuning capacitor is located
within this box, as seen in figure 6. An external bias supply is required for proper amplifier operation. Operating voltages may be determined from the instruction sheets for the
particular tube to be employed.
Whenever the amplifier enclosure requires
panel door for coil changing access it is wise
to place a power interlock on the door that
will turn off the high voltage supply whenever
the door is open!
a
THE RADIO
H.F. Power Amplifiers
Figure
5
LEFT -HAND VIEW OF KILOWATT
AMPLIFIER OF FIGURE 4
Above shielded meter box is the protective
"micro- switch" which opens the primary power
circuit when the panel door is not closed. Tube
sockets are recessed in the chassis so that
top of tube socket shells are about 1,2-inch
above chassis level. On right side of amplifier
(facing it from the rear) the tube socket is
nearest the panel, with the neutralizing capacitor behind it. On the opposite side, the
capacitor is nearest the panel with the tube
directly behind it. This layout transposition
produces very short neutralizing leads, since
connections may be made through the stator
of plate tuning capacitor.
29 -3
Push -Pull
Tetrode Amplifiers
Tetrode tubes may be employed in push -pull
amplifiers, although the modern trend is to
parallel operation of these tubes. A typical
circuit for push -pull operation is shown in
figure 7. The remarks concerning the filament
supply, plate feed, and grid bias in Section
29 -2 apply equally to tetrode stages. Because
of the high circuit gain of the tetrode amplifier, extreme care must be taken to limit
interstage feedback to an absolute minimum.
Many amateurs have had bad luck with tetrode tubes and have been plagued with parasitics and spurious oscillations. It must be remembered with high gain tubes of this type
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HANDBOOT(
P -P
Tetrode Amplifier
607
Figure 7
Figure 6
UNDER CHASSIS VIEW OF
1- KILOWATT TRIODE AMPLIFIER
The grid circuit tuning capacitor and plate
circuit r -f choke are contained in the below
chassis enclosure formed by a small chassis
mounted at right angles to the front panel.
The bondswitch coil assembly for the grid
circuit is mounted on two brackets above this
cutout. A metal screen attached to the bottom
of the amplifier completes the TVI -proof
enclosure.
that almost full output can be obtained with
practically zero grid excitation. Any minute
amount of energy fed back from the plate circuit to the grid circuit can cause instability or
oscillation. Unless suitable precautions are incorporated in the electrical and mechanical design of the amplifier, this energy feedback will
inevitably occur.
Fortunately these precautions are simple.
The grid and filament circuits must be isolated
from the plate circuit. This is done by placing
these circuits in an "electrically tight" box.
All leads departing from this box are by -passed and filtered so that no r -f energy can pass
along the leads into the box. This restricts the
energy leakage path between the plate and grid
circuits to the residual plate -to -grid capacity
of the tetrode tubes. This capacity is of the
order of 0.25 µpfd. per tube, and under normal
conditions is sufficient to produce a highly regenerative condition in the amplifier. Whether
or not the amplifier will actually break into oscillation is dependent upon circuit losses and
residual lead inductance of the stage. Suffice
to say that unless the tubes are actually neutralized a condition exists that will lead to
circuit instability and oscillation under certain operating conditions. With luck, and a
CONVENTIONAL PUSH -PULL
TETRODE AMPLIFIER CIRCUIT
Push -pull amplifier
uses many of the same
required by triode tubes (see
figure 2). Screen supply is also required.
Blower for filament seals of tubes.
C. -Low internal inductance capacitor, .001
pfd., SKV. Centralab type 858S -1000.
NC -See text and figure 8.
PC- Parasitic choke. 50 ohm, 2 -watt composition resistor wound with 3 turns =12 e.
components
B-
wire.
Note: Strap multiple screen terminals together
at socket with Je" copper ribbon. Attach
PC to center of strap.
heavily loaded plate circuit, one might be able
to use an un- neutralized push -pull tetrode amplifier stage and suffer no ill effects from the
residual grid -plate feedback of the tubes. In
fact, a minute amount of external feedback in
the power leads to the amplifier may just (by
chance) cancel out the inherent feedback of
the amplifier circuit. Such a condition, however, results in an amplifier that is not "reproduceable." There is no guarantee that a duplicate amplifier will perform in the same, stable manner. This is the one, great reason that
many amateurs having built a tetrode amplifier
that "looks just like the one in the book" find
out to their sorrow that it does not "work like
the one in the book."
This borderline situation can easily be overcome by the simple process of neutralizing the
high -gain tetrode tubes. Once this is done, and
the amplifier is tested for parasitic oscillations (and the oscillations eliminated if they
occur) the tetrode amplifier will perform in an
excellent manner on all bands. In a word, it
will be "reproduceable."
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608
1
THE RADIO
H.F. Power Amplifiers
-
,,,
.ita....
.:
1
1
Figure 9
UNDER CHASSIS VIEW OF
4 -250A AMPLIFIER
Figure 8
REAR VIEW OF PUSH PULL
4 -250A AMPLIFIER
The neutralizing rods are mounted on ceramic
feedthrough insulators adjacent to each tube
socket. Low voltage power leads leave the
grid circuit compartment via Hypass capacitors located on the lower left corner of the
chassis. A screen plate covers the rear of the
amplifier during operation. This plate was
removed for the photograph.
The bias supply for the amplifier is mounted
at the front of the chassis between the two
control shafts. A blower motor is mounted
beneath each tube socket. A screened plate
is placed on the bottom of the chassis to complete the under -chassis shielding.
-
As a summation, three requirements must
be met for proper operation of tetrode tubes
whether in a push-pull or parallel mode:
1. Complete isolation must be achieved between the grid and plate circuits.
2. The tubes must be neutralized.
3. The circuit must be parasitic -free.
Amplifier
Construction
The push -pull tetrode ampli fier should be built around
two "r -f tight" boxes for the
grid and plate circuits. A typical layout that has
proven very satisfactory is shown in figures 8
and 9. The amplifier is designed around a Barker & Williamson "butterfly" tuning capacitor.
The 4 -250A tetrode tubes are mounted at the
rear of the chassis on each side of the capacitor. The base shells of the tubes are grounded
by spring clips, and short adjustable rods project up beside each tube to act as neutralizing
capacitors. The leads to these rods are cross connected beneath the chassis and the rods
provide a small value of capacitance to the
plates of the tubes. This neutralization is necessary when the tube is operated with high
power gain and high screen voltage. As the
operating frequency of the tube is increased,
the inductance of the internal screen support
lead of the tube becomes an important part of
the screen ground return circuit. At some critical frequency (about 45 Mc. for the 4 -250A
tube) the screen lead inductance causes a
series resonant condition and the tube is said
to be "self- neutralized" at this frequency.
Above this frequency the screen of the tetrode
tube cannot be held at ground potential by the
usual screen by -pass capacitors. With normal
circuitry, the tetrode tube will have a tendency
to self -oscillate somewhere in the 120 Mc. to
160 Mc. region. Low capacity tetrodes that can
operate efficiently at such a high frequency are
capable of generating robust parasitic oscillations in this region while the operator is vainly trying to get them operating at some lower
frequency. The solution is to introduce enough
loss in the circuit at the frequency of the
parasitic so as to render oscillation impossible. This procedure has been followed in this
amplifier.
During a long series of experiments designed to stabilize large tetrode tubes, it was
found that suppression circuits were most effective when inserted in the screen lead of the
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HANDBOOK
Pi- Network Amplifiers
tetrode. The screen, it seemed, would have r -f
potentials measuring into the thousands of volts
upon it during a period of parasitic oscillation.
By- passing the screen to ground with copper
strap connections and multiple by-pass capacitors did little to decrease the amplitude of the
oscillation. Excellent parasitic suppression was
brought about by strapping the screen leads
of the 4 -250A socket together (figure 7) and
inserting a parasitic choke between the screen
terminal of the socket and the screen by -pass
capacitor.
After this was done, a very minor tendency
towards self -oscillation was noted at extremely high plate voltages. A small parasitic choke
in each grid lead of the 4 -250A tubes eliminated this completely.
The neutralizing rods are mounted upon two
feedthrough insulators and cross -connected to
the 4 -250A control grids beneath the chassis.
These rods are threaded so that they may be
run up and down the insulator bolt for neutralizing adjustment.
Because of the compact size of many tetrodes
it is necessary to cool the filament seals of the
tube with a blast of air. A small blower can be
mounted beneath the chassis to project cooling
air directly at the socket of the tube as shown
in figure 9.
Inductive Tuning of
Push -Pull Amplifiers
The plate tank circuit
of the push -pull amplifier must have a low
impedance to ground at harmonic frequencies
to provide adequate harmonic suppression. The
usual split- stator tank capacitor, however, has
an uncommonly high impedance in the VHF
region wherein the interference -causing harmonics lie. A push -pull vacuum-type capacitor
may be used as these units have very low internal inductance, but the cost of such a capac-
609
seen in figure 10. Two fixed vacuum capacitors are mounted vertically upon the chassis
and the upper terminals are attached to the
plates of the amplifier tubes by means of low
impedance straps. Resonance is established by
rotation of a shorted copper loop located within the amplifier tank coil. This loop is made
of a %8" long section of copper water pipe,
two inches in diameter. Approximate resonance
is established by varying the spacing between
the turns of the copper tubing tank coil. Inductive coupling is used between the tank coil
and the antenna circuit in the usual manner.
Sufficient range to enable the operator to cover
a complete high frequency band may be had
with this interesting tuning method.
29 -4
Tetrode PiNetwork Amplifiers
The most popular amplifier today for both
commercial and amateur use is the pi- network
configuration shown in figure 11. This circuit
is especially suited to tetrode tubes, although
triode tubes may be used under certain circumstances.
A common form of pi- network amplifier is
shown in figure 11A. The pi circuit forms the
matching system between the plate of the amplifier tube and the low impedance, unbalanced
antenna circuit. The coil and input capacitor
itor is quite high.
A novel solution to this problem is to employ a split stator capacitor made up of two
inexpensive fixed vacuum capacitors. Amplifier adjustment can then best be accomplished
by inductive tuning of the plate tank coil as
Figure 10
INDUCTIVE TUNING MAY BE
EMPLOYED IN HIGH POWER
AMPLIFIER
Two fixed vacuum capacitors form split -stator
capacitance, providing very low inductance
ground path for plate circuit harmonics. Tuning is accomplished by means of shorted,
single -turn link placed in center of tank
coil. Shorted link is made from 3k -inch section
cut from copper water pipe. Larger link outside of tank coil is antenna pick -up coil.
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610
H.F.
THE RADIO
Power Amplifiers
LOW Z
OUTPUT
EXCITATION
- BIA'
15V.1,
Figure 11
TYPICAL PI- NETWORK CONFIGURATIONS
circuit provides out -of -phase voltage for grid neutralization of tetrode tube. Rotary coil
employed in plate circuit, with small, fixed auxiliary coil for 28 Mc. Multiple tuning grid tank TI
covers 3.5 - 30 Mc. without switching.
Tapped grid and plate inductors are used with "bridge type" neutralizing circuit for tetrode amplifier stage. Vacuum tuning capacitor is used in input section of pi- network.
Untuned input circuit (resistance loaded) and plate inductor ganged with tuning capacitor comprise
simple amplifier configuration.
PCr, PC.-57 ohm, 2 watt composition resistor, wound with 3 turns o 18 c. wire.
A -Split grid
is
BC-
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HANDBOOK
Pi- Network
of the pi may be varied to tune the circuit over
a 10 to 1 frequency range (usually 3.0 - 30
Mc.) . Operation over the 20 - 30 Mc. range
takes place when the variable slider on coil L2
is adjusted to short this coil out of the circuit.
Coil L. therefore comprises the tank inductance
for the highest portion of the operating range.
This coil has no taps or sliders and is constructed for the highest possible Q at the high
frequency end of the range. The adjustable
coil ( because of the variable tap and physical
construction) usually has a lower Q than that
of the fixed coil.
The degree of loading is controlled by capacitors C. and G. The amount of circuit capacity required at this point is inversely proportional to the operating frequency and to
the impedance of the antenna circuit. A loading capacitor range of 100 µµfd. to 2500
µµfd. is normally ample to cover the 3.5 - 30
Mc. range.
The pi circuit is usually shunt-fed to remove
the d.c. plate voltage from the coils and capacitors. The components are held at ground potential by completing the circuit ground
through the choke RFC,. Great stress is placed
upon the plate circuit choke RFC:. This component must be specially designed for this
mode of operation, having low inter -turn capacity and no spurious internal resonances
throughout the operating range of the amplifier.
Parasitic suppression is accomplished by
means of chokes PC -I and PC -2 in the screen
and grid leads of the tetrode. Suitable values
for these chokes are given in the parts list of
figure 1L Effective parasitic suppression is dependent to a large degree upon the choice of
screen bypass capacitor C2. This component
must have extremely low inductance throughout the operating range of the amplifier and
well up into the VHF parasitic range. The capacitor must have a voltage rating equal to at
least twice the screen potential (four times the
screen potential for plate modulation). There
are practically no capacitors available that will
perform this difficult task. One satisfactory solution is to allow the amplifier chassis to form
one plate of the screen capacitor. A "sandwich"
is built upon the chassis with a sheet of insulating material of high dielectric constant and
a matching metal sheet which forms the screen
side of the capacitance. A capacitor of this
type has very low internal inductance but is
very bulky and takes up valuable space beneath the chassis. One suitable capacitor for
this position is the Centralab type 858S-1000,
Amplifiers
611
BIAS SUPPLY
SCREEN SUPPLY
P= PLATE SUPPLY
B
=
S
OPERATE
TUNE
"COMMON
MINUS"
LEAD
BY-
Figure 12
GROUNDED SCREEN GRID
CONFIGURATION PROVIDES HIGH
ORDER OF ISOLATION IN TETRODE
AMPLIFIER STAGE
A- Typical amplifier circuit has cathode return at ground potential. All circuits return to cathode.
-All
circuits return to cathode, but ground
point has been shifted to screen terminal
of tube. Operation of the circuit remains
the same, as potential differences between
elements of the tube are the same as in
circuit A.
C- Practical grounded screen circuit. "Common minus" lead returns to negative of
plate supply, which cannot be grounded.
Switch S: removes screen voltage for tuneB
up purposes.
rated at 1000 µµfd. at 5000 volts. This compact ceramic capacitor has relatively low internal inductance and may be mounted to the
chassis by a 6 -32 bolt. It is shown in various
amplifiers described in this chapter. Further
screen isolation may be provided by a shielded
power lead, isolated from the screen by a .001
pfd. ceramic capacitor and a 100 ohm carbon resistor.
Various forms of the basic pi- network amplifier are shown in figure 11. The A configuration employs the so- called "all- band" grid
tank circuit and a rotary pi- network coil in the
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612
THE RADIO
H.F. Power Amplifiers
plate circuit. The B circuit uses coil switching
in the grid circuit, bridge neutralization, and
a tapped pi- network coil with a vacuum tuning capacitor. Figure 1 1C shows an interesting
circuit that is becoming more popular for
class ABI linear operation. A tetrode tube operating under class ABI conditions draws no
grid current and requires no grid driving
power. Only r -f voltage is required for proper
operation. It is possible therefore to dispense
with the usual tuned grid circuit and neutralizing capacitor and in their place employ a simple load resistor in the grid circuit across
which the required excitation voltage may be
developed. This resistor can be of the order of
50 - 300 ohms, depending upon circuit requirements. Considerable power must be dissipated in the resistor to develop sufficicnt
grid swing, but driving power is often cheaper
to obtain than the cost of the usual grid circuit components. In addition, the low impedance grid return removes the tendency towards instability that is so common to the
circuits of figure 1IA and 11B. Neutralization
is not required of the circuit of figure 11C,
and in many cases parasitic suppression may
be omitted. The price that must be paid is the
additional excitation that is required to develop operating voltage across grid resistor R..
The pi- network circuit of figure 11C is
interesting in that the rotary coil L. and the
plate tuning capacitor Ca are ganged together
by a gear train, enabling the circuit to be tuned
to resonance with one panel control instead of
the two required by the circuit of figure 11A.
Careful design of the rotary inductor will permit the elimination of the auxiliary high frequency coil L,, reducing the cost and complexity of the circuit.
result of this loss of circuit isolation. A solution to this problem is to eliminate the screen
bypass capacitor, grounding the screen terminals of the tube by means of a low inductance strap. Screen voltage is then applied to
the tube by grounding the positive terminal
of the screen supply, and "floating" the negative of the screen and bias supplies below
ground potential as shown in figure 12.
Meters are placed in the separate circuit
cathode return leads, and each meter reads
only the current flowing in that particular
circuit. Operation of this grounded screen circuit is normal in all respects, and it may be
applied to any form of grid- driven tetrode
amplifier with good results.
29 -5
Grounded -Grid
Amplifier Design
The grounded grid (g -g) amplifier has
achieved astounding popularity in recent years
as a high power linear stage for sideband application. Various versions of this circuit are
illustrated in figure 13. In the basic circuit,
the control grid of the tube is at r.f. ground
potential and the exciting signal is applied to
the cathode by means of a tuned circuit. Since
the grid of the tube is grounded, it serves as
a shield between the input and output circuits, making neutralization unnecessary in
many instances. The very small plate to cathode
capacitance of most tubes permits a minimum
of intrastage coupling below 30 Mc. In addition, when zero bias triodes or tetrodes are
used, screen or bias supplies are not usually
required.
Configuration
The Grounded Screen For maximum shielding,
it is necessary to operate
Feedthrough Power
the tetrode tube with the
screen at r.f. ground potential. As the screen
has a d.c. potential applied to it (in grid driven circuits), it must be bypassed to ground
to provide the necessary r.f. return. The bypass capacitor employed must perform efficiently over a vast frequency spectrum that
includes the operating range, plus the region
of possible v.h.f. parasitic oscillations. This is
a large order, and the usual bypass capacitors
possess sufficient inductance to introduce regeneration into the screen circuit, degrading
the grid -plate shielding to a marked degree.
Nonlinearity and self -oscillation can be the
power appears in the plate
circuit of the grounded grid (cathode driven)
amplifier and is termed feedthrough power.
In any amplifier of this type, whether it be
triode or tetrode, it is desirable to have a
large ratio of feedthrough power to peak grid
driving power. The feedthrough power acts
as a swamping resistor across the driving circuit to stabilize the effects of grid loading.
The ratio of feedthrough power to driving
power should be about 10 to 1 for best stage
linearity. The feedthrough power provides the
user with added output power he would not
obtain from a more conventional circuit. The
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A portion of the exciting
HANDBOOK
Grounded -Grid Amplifier
R.r. OUT
613
R.r. OUT
EXC
E
xC.
R.r. OUT
EXC.
©
Figure 13
THE GROUNDED-GRID AMPLIFIER
linear amplifier for sideband service, the grounded -grid circuit provides economy
Widely used as a
and simplicity, in addition to a worthwhile reduction in intermodulation distortion. A -The basic
g -g amplifier employs tuned input circuit. B
simplified circuit employs untuned r.f. choke in
cathode in place of the tuned circuit. Linearity and power output are inferior compared to circuit
of figure A.
Simple high -C pi- network may be used to match output impedance of sideband
exciter to input impedance of grounded-grid stage.
Parallel- tuned, High -C circuit may be
employed for bandswitching amplifier. Excitation tap is adjusted to provide low value of sm.,. on
exciter coaxial line.
-A
C-
D-
driver stage for the grounded grid amplifier
must, of cous-se, supply the normal excitation
power plus the feedthrough power. Many commercial sideband exciters have power output
capabilities of the order of 70 to 100 watts
and are thus well suited to drive high power
grounded grid linear amplifier stages whose
total excitation requirements fall within this
range.
Distortion
Laboratory measurements made on
various tubes in the circuit of
figure 13A show that a distortion
reduction of the order of 5 to 10 decibels in
odd -order products can be obtained by operating the tube in grounded grid service as opposed to grid- driven service. The improvement
in distortion varies from tube type to tube
type, but some order of improvement is noted
for all tube types tested. Most amateur -type
transmitting tubes provide signal -to- distortion
ratios of -20 to -30 decibels at full output
in class ABI grid- driven operation. The ratio
increases to approximately -25 to -40 deciProducts
bels for class B grounded grid operation. Dis-
tortion improvement is substantial, but not as
great as might otherwise be assumed from
the large amount of feedback inherent in the
grounded grid circuit.
A simplified version of the grounded grid
amplifier is shown in figure 13B. This configuration utilizes an untuned input circuit,
and is very popular as an inexpensive and
simplified form of the more sophisticated circuit of figure 13A. It has inherent limitations,
however, that should be recognized. In general, slightly less power output and efficiency
is observed with the untuned cathode circuit,
odd -order distortion products run 4 to 6
decibels higher, and the circuit is harder to
drive and match to the exciter than is the
tuned cathode circuit of figure 13A. For maximum linearity and optimum operation, a certain amount of "flywheel" effect is required in
the cathode input that can only be supplied
by a high -C tuned circuit of some form.
Since the single ended class B grounded
grid linear amplifier draws grid current on
only one -half (or less) of the operating cycle,
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THE RADIO
H.F. Power Amplifiers
-BIAS
R.F.OUT.
EXC.
Figure 14
TETRODE TUBES MAY BE USED IN GROUNDED-GRID AMPLIFIERS
Tetrodo tube may be used in cathode driven configuration, with bias and screen
voltages applied to elements which are at r.f. ground potential. B-Grid current of
grounded -grid tube is easily monitored by R -C network which lifts grid above ground
sufficiently to permit a millivoltmeter to indicate voltage drop across -ohm resistor.
Meter is a 0 -1 d.c. milliameter in series with appropriate multiplier resistor.
A-
1
the sideband exciter "sees" a low impedance
load during this time, and a very high impedance load over the balance of the cycle.
Linearity of the exciter is thereby affected and
the distortion products of the exciter are enhanced. Thus, the driving signal is degraded
in the cathode circuit of the grounded grid
stage unless the unbalanced input impedance
can be modified in some fashion. A high -C
tuned circuit stores enough energy over the
operating r.f. cycle so that the exciter "sees"
a relatively constant load at all times. In addition, the tuned circuit may be tapped or otherwise adjusted so that the standing wave ratio
on the coaxial line coupling the exciter to the
amplifier is relatively low. This is a great
advantage, particularly in the case of those
exciters having fixed -ratio pi- network output
circuits designed expressly for a 50 -ohm
termination.
Finally, it must be noted that removal of
the tuned cathode circuit breaks the amplifier
plate circuit return to the cathode, and r.f.
plate current pulses must return to the cathode
via the outer shield of the driver coaxial line
and back via the center conductor! Extreme
fluctuations in exciter loading, intermodulation distortion, and TVI can be noticed by
changing the length of the cable between the
exciter and the grounded grid amplifier when
an untuned cathode input circuit is employed.
Design features of the single ended and push -pull ampliConstruction
fiers discussed previously
apply equally well to the
grounded grid stage. The g -g linear amplifier
Grounded Grid
Amplifier
may have either configuration, although the
majority of g -g stages are single- ended, as
push -pull offers no distinct advantages and
adds greatly to circuit complexity.
The cathode circuit of the amplifier is
resonated to the operating frequency by means
of a high -C tank (figure 13A). Resonance is
indicated by maximum grid current of the
stage. A low value of s.w.r. on the driver coaxial line may be achieved by adjusting the
tap on the tuned circuit, or by varying the
capacitors of the pi- network (figure 13C).
Correct adjustments will produce minimum
s.w.r. and maximum amplifier grid current at
the same settings. The cathode tank should
have a Q of 2 or more.
The cathode circuit should be completely
shielded from the plate circuit. It is common
practice to mount the cathode components in
an "r.f. tight" box below the chassis of the
amplifier, and to place the plate circuit components in a screened box above the chassis.
The grid (or screen) circuit of the tube is
operated at r.f. ground potential, or may have
d.c. voltage applied to it to determine the
operating parameters of the stage (figure
14A ). In either case, the r.f. path to ground
must be short, and have extremely low inductance, otherwise the screening action of the
element will be impaired. The grid (and
screen) therefore, must be bypassed to ground
over a frequency range that includes the operating spectrum as well as the region of possible
v.h.f. parasitic oscillations. This is quite a
large order. The inherent inductance of the
usual bypass capacitor plus the length of
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HANDBOOK
Grounded -Grid Amplifier
element lead within the tube is often sufficient to introduce enough regeneration into
the circuit to degrade the linearity of the amplifier at high signal levels even though the
instability is not great enough to cause parasitic oscillation. In addition, it is often desired
to "unground" the grounded screen or grid
sufficiently to permit a metering circuit to
be inserted.
One practical solution to these problems is
to shunt the tube element to ground by means
of a 1 -ohm composition resistor, bypassed with
a .01 µfd. ceramic disc capacitor. The voltage
drop caused by the flow of grid (or screen)
current through the resistor can easily be
measured by a milli -voltmeter whose scale
is calibrated in terms of element current
(figure 14B).
The plate circuit of the grounded grid amplifier is conventional, and either pi- network
or inductive coupling to the load may be used.
There is some evidence to support the belief
that intermodulation distortion products are
reduced by employing plate circuit Q's somewhat higher than normally used in class -C
amplifier design. A circuit Q of 15 or greater
is thus recommended
for grounded grid
amplifier plate circuits.
Tuning the
Since the input and output
circuits of the grounded grid
Amplifier
amplifier are in series, a certain proportion of driving
power appears in the output circuit. If full
excitation is applied to the stage and the output circuit is opened, or the plate voltage removed from the tube, practically all of the
driving power will be dissipated by the grid
of the tube. Overheating of this element will
quickly occur under these circumstances, followed by damage to the tube. Full excitation
should therefore never be applied to a
grounded grid stage unless plate voltage is
applied beforehand, and the stage is loaded
to the antenna.
For best linearity, the output circuit of the
grounded grid stage should be overcoupled
so that power output drops about 2- percent
from maximum value. A simple output r.f.
voltmeter is indispensable for proper circuit
adjustment. Excessive grid current is a sign of
antenna undercoupling, and overcoupling is
indicated by a rapid drop in output power.
Proper grounded grid stage operation can be
Grounded Grid
615
determined by finding the optimum ratio between grid and plate current and by adjusting
the drive level and loading to maintain this
ratio. Many manufacturers now provide
grounded grid operation data for their tubes,
and the ratio of grid to plate current can be
determined from the data for each particular
tube.
Not all tubes are suitable
for grounded grid service. In
addition, the signal- to -disdistortion ratio of the suitable tubes varies
over a wide range. Some of the best g -g performers are the 811A, 813, 7094, 4 -125A,
4-250A, 4 -400A and 4- 1000A. In addition,
the 3 -400Z and 3 -1000Z triodes are specifically designed for low distortion, grounded
grid amplifier service. The older types 837
and 803 are used extensively for g -g operation
but are not recommended because of poor
signal -to- distortion ratios.
Certain types of tetrodes, exemplified by the
4 -65A, 4X150A, 4CX300A and 4CX1000A
should not be used as grounded grid amplifiers unless grid bias and screen voltage are
applied to the elements of the tube (figure
14A). The internal structure of these tubes
permits unusually high values of grid current
to flow when true grounded grid circuitry is
used, and the tube may be easily damaged by
this mode of operation.
The efficiency of a typical grounded grid
amplifier runs between 55- and 65-percent,
indicating that the tube employed should have
plenty of plate dissipation. In general, the
p.e.p. input in watts to a tube operating in
grounded grid configuration can safely be
about 2.5 to 3 times the rated plate dissipaChoice of Tubes
for G -G Service
tion. Because of the relatively low average -topeak power of the human voice it is tempting
to push this ratio to a higher figure in order
to obtain more output from a given tube.
This action is unwise in that the odd -order
distortion products rise rapidly when the tube
is overloaded, and because no safety margin is
left for tuning errors or circuit adjustments.
Neutralization
At some high frequency the
shielding action of the grid
of the g -g amplifier deteriorates. Neutralization may be necessary at
higher frequencies either because of the presence of inductance between the active grid
of the
G -G Stage
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616
THE RADIO
H.F. Power Amplifiers
n. r. ouT.
EXC.
O
Figure 15
NEUTRALIZING CIRCUITS FOR GROUNDED -GRID STAGES
Neutralization of the g -g stage may be necessary at the higher frequencies. Energy fed
back in proper phase from plate to cathode is used to neutralize the unwanted energy
fed through the tube (A). Reactance placed in series with the grid return lead (B) will
accomplish the same result. The inductance L usually consists of the internal grid lead
of the tube, and capacitor C may be the grid bypass capacitor. A series resonant circuit
at the operating frequency is thus formed.
element and the common returns of the input
and output circuit, or because of excessive
plate- cathode capacitance.
Neutralization, where required, may be accomplished by feeding out -of -phase energy
from the plate circuit to the filament circuit
(figure 15A) or by inserting a reactance in
series with the grid (figure 15B). For values
of plate- cathode capacitance normally encountered in tubes usable in g -g service, the residual inductance in the grid -ground path provides sufficient reactance, and in some cases
even series capacitance will be required. Typical
tube electrode capacitances are shown in figure
16A. These can be represented by an equiva-
lent star connection of three capacitors (figure
16B). If an inductance L is placed in series
with C, so that a resonant circuit is formed
(figure 16C), point O will be at ground
potential (figure 16D) This will prevent the
transfer of energy from point P to point K,
since there now exists no common coupling
impedance. The determination of the value of
C, and L are shown in the drawing.
It is apparent that when the plate- cathode
capacitance of the tube is small as compared
to the plate -grid and the grid- cathode capacitances, C, is a large value and the required
value of inductance L is small. In practical
cases the value of L is supplied by the tube
.
Figure 16
Tube electrode capacitances can be represented by an equivalent star connection of three capacitors. If inductance is placed in series with C, so that a resonant circuit is formed (drawing C),
point 0 will be at ground potential.
O
CP-K
CC °
CG-P
t
CP-K
X
CG-K
CG-K
X
C-G
C-K
t
L
(2Rf)2 X
Cc
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HANDBOOK
350 Watt P.E.P. Amplifier
and lead inductance, and the grid to ground
impedance can be closely adjusted by proper
choice of the bias bypass capacitor (figure
15B). Below a certain frequency determined
by the physical geometry of the tube, neutralization may be accomplished by adding inductance to the grid return lead; above this
frequency it may be necessary to series tune
the circuit for minimum energy feedthrough
from cathode to plate. Most tubes are sufficiently well screened so that series inductive
neutralization at the lower frequencies is unnecessary, but series capacitance tuning of the
grid return lead may be required to prevent
oscillation at some parasitic frequency in the
v.h.f. range.
29 -6 A 350 Watt P.E.P.
Grounded -Grid
Amplifier
This section features an extremely stable,
five band, grounded grid linear amplifier for
sideband service. Employing the 7094 beam
power tube, the amplifier provides band switched operation on all bands between 80
and 10 meters. Power output is in excess of
200 watts, and third order distortion products
are better than -30 decibels below maximum
two -tone signal level.
High power gain, high efficiency, and low
distortion can be provided economically by a
high -s triode tube operating in grounded grid
configuration. Beam power tubes or tetrodes
(such as the 7094, 813 or 4 -250A) which
can be operated as high -µ triodes make excellent grounded grid amplifiers. As a class B
linear amplifier in sideband service, a triode connected 7094 with forced air cooling of the
envelope can handle a conservative peak envelope -power input (p.e.p.) of 350 watts
with only I750 volts on the plate and zero
bias on the grids. For full input, a sideband
exciter capable of an output of only 15 watts
p.e.p. is required.
The amplifier, complete with power supply,
fits on a standard 101/2 -inch relay rack panel
which may be placed within a cabinet for use
directly on the operating table.
Amplifier
Circuit
617
The circuit of the amplifier and
power supply is shown in figure
17. The plate output circuit is a
bandswitching pi- network using two tapped
coils and a shorting switch. The position of the
taps are chosen to provide an operating Q of
15 or better on all bands with a 50 -ohm
antenna load. An auxiliary loading capacitor
is switched into the circuit in the 80 meter
position of the bandswitch. For low impedance
antennas (below 50 ohms) this capacitor
should be increased in value to 1000 ppfd.
The grid and screen of the 7094 tube are
at r.f. ground potential. The d.c. screen return
is to the cathode of the tube, and the panel
meter (M1) is switched so that it is possible to read either grid current or plate current. The meter is a single- scale, 0 -300 d.c.
milliammeter. A lower range meter and external shunt were not considered necessary
because the normal peak grid current (80 ma.)
and peak plate current (200 ma.) can easily
be read on the same scale. A 1000 -ohm resistor is connected between the positive terminal of the meter and ground to prevent
high voltage from appearing at the cathode of
the tube in the event of switch failure.
An untuned input circuit is used in the
cathode for simplicity. An alternative tuned
input circuit is shown. Use of the tuned circuit will result in better linearity and lower
driving power requirements. If the tuned circuit is omitted, it may be necessary to "prune"
the coaxial line between the exciter and the
amplifier to achieve maximum driving voltage
in the cathode circuit. A circuit Q of two or
more is required in this tank.
The power supply is a conventional full
wave circuit with a choke input filter. Type
3B28 gas rectifier tubes are used in place of
866A's to eliminate the "hash" produced by
the mercury vapor tubes and to permit the
amplifier to be operated on its side during
tests and measurements. 866A's may be used
in place of the 3B28's without any circuit
changes provided the amplifier is always positioned so that the tubes are vertical.
The plate switch is connected in series with
the filament switch so that plate power cannot
be applied to the rectifier tubes until the
filament circuit is energized. Filaments should
be allowed to warm up for 30 seconds before
plate voltage is turned on.
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THE RADIO
H.F. Power Amplifiers
618
Amplifier
Because of the simplicity of the
circuit it is possible to construct the amplifier and power
supply on a single 12" x 17" x 3" aluminum
chassis. The chassis is attached to a 101/2"
relay rack panel by means of two chassis
mounting brackets. The 7094 and plate tank
circuit components are enclosed in a 7" x 12"
x 91A " box made of 18 -gauge sheet aluminum.
The front of the box is mounted flush against
the rack panel, and both are drilled simulConstruction
taneously for the shafts of the plate tuning
and loading capacitors and the bandswitch.
Half -inch wide flanges on the top and bottom
of the enclosure provide good r.f. contact to
the chassis and to the perforated aluminum
cover plate.
The small fan mounted on the rear wall of
the box provides forced -air cooling for the
7094. The air intake hole is 3- inches in diameter and covered with perforated aluminum
stock.
Figure 17
SCHEMATIC, GROUNDED -GRID AMPLIFIER
C1
-100
;LAM.,
3
kv.
Johnson
100E30
(#155-10)
C2-500
µpfd., 2 kv. Johnson 500E20
(#154-3)
C3 -See text. 500 ppfd. mica, 1250 volts
C4
section b.c. capacitor, 1100 ppld.
Miller 2113
-3
-8H,
250 ma. Thordarson 20056
amp. fuse, size 3AG
amp. fuse, size 3AG "slo -blo"
L1 -10 and 15 meter coil: 9 turns of 3/16 inch copper tubing, 2" inside diameter, 1/2inch spacing between turns. 10 meter tap is
4/2 turns from plate end al coil. 15 meter
tap is at junction between L1 and L2
L2-23 turns, B&W #3095 -1 inductor. 20 and
40 meter taps are 19 and 10 turns, respectively, from the output end of coil. Number
12 wire, 2/2" diameter
L3 -18 turns # 16 wire, 1" diam., 3" long, 6
turns per inch (Air -Dux #806-T), 2.3 ph.
CH1
-5
F2 -1
F1
40 meter tap (1 ph) at 9 turns, 20 meter
top (0.5 ph) at 41/2 turns, I5 meter tap
(0.3 ph) at 3 turns, 10 meter tap (0.15 ph)
at 11/2 turns. All taps measured from
ground end of coil
P1,P2 -115 volt pilot lamp assembly
PC
turn of 1/2 -inch plate strap, 1/2-inch
diameter wound about three 100 ohms, 2
watt composition resistors in parallel
RFC1 -2.5
mh, 300 ma. National R -300,
placed between pins 4 and 7 of tube socket
RFC2 -0.225 mh, 800 ma. National R -175A
RFC3 -2.5 mh, 100 ma. National R -100
Single pole, 5 position ceramic switch.
Ohmite #111 or equivalent
T,-6.3 volt @ 4 amp. Stancor P-4019
T2 -2.5 volt @ 10 a. Thordarson 21F02
T3-2065 -0 -2065 volts @ 200 ma. (1750 v.,
d.c.) Stancor PT-8315
Blowers- Cooling motor and fan. Shaded -pole
induction motor, 2400 r.p.m. with 4- bladed
fan, 21/2" diem. Allied Radio Co., Chicago,
Ill. Part number 72P -715
-1
SI-
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HANDBOOK
350 Watt
P.E.P.
Amplifier
619
Figure 18A
350 WATT P.E.P.
AMPLIFIER AND
POWER SUPPLY
the 7094 beam
tube, this compact, grounded-grid amplifier may be driven to
full input with a 15 -watt
The
exciter.
sideband
complete amplifier and
power supply mount behind a 101/2" relay rack
panel. Panel controls are
(I. to r.): meter switch,
plate tuning (above) and
filament switch (below),
Using
power
bandswitch,ontennaloading (above) and plate
switch (below).
Figure 18B
REAR VIEW OF
AMPLIFIER
The power supply
ponents
are
com-
grouped
of the
chassis. R.f. input receptacle and 115 -volt power
receptacle are placed on
rear apron of chassis.
Antenna receptacle is
mounted on rear wall of
shielded enclosure. Ceramic disc capacitor is
placed across meter leads
directly at terminals, and
leads are run in shielded
braid to under- chassis
area.
about
one
end
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620
H.F. Power Amplifiers
To hold r.f. loss to a minimum, connections
between the plate tank circuit components are
made of 1/4 -inch wide silver plated copper
strap. Your local jeweler can probably handle
the silver plating job for you.
A short length of RG -8 /U coaxial line is
used to make the connection between the loading capacitor and the coaxial antenna receptacle located on the rear of the enclosure. The
Figure 19
TOP VIEW OF LINEAR AMPLIFIER
The plate circuit is enclosed in r.f. -tight compartment bolted to the chassis deck. Power
supply choke is outside rear of compartment,
with plate transformer and 3B28 rectifier
tubes to the right. Enclosure is covered with
a piece of perforated aluminum plate for
maximum ventilation. The 7094 tube is at
center of enclosure, with 10 -15 meter coil between it and bandswitch. Loading capacitor is
to the left, with the 20 -80 meter coil directly
above it. Plate capacitor and r.f. choke are
at the right.
T H
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outer braid of the line is grounded at one end
to the frame of the capacitor and at the other
end to the shell of the coaxial receptacle.
A single 8 pfd. filter capacitor is too large
to fit beneath the chassis, so four 2 pfd. units
are wired in parallel to provide sufficient
capacity for good dynamic regulation. These
capacitors, together with the filament transformers and bleeder resistors are placed in a
free corner of the under -chassis area.
Amplifier Tuning
and Adjustment
All wiring should be
checked before power is
applied to the amplifier.
The d.c. resistance to ground of the B -plus
line should be about 100,000 ohms. The amplifier is connected to the exciter and to the
antenna or to a 200 watt, 50 -ohm dummy
load. Mesh the plates of the loading capacitor
and place the meter switch in the plate current
www.americanradiohistory.com
HANDBOOK
350 Watt P.E.P. Amplifier
position. Turn on the plate voltage and note
the resting plate current. It should be about
35 to 40 milliamperes. Apply a low level single
tone signal (carrier) to the amplifier and tune
the plate tank circuit to resonance. Switch the
meter to indicate grid current, and advance
the excitation level until the grid current reading is about 50 milliamperes. Reduce the loading capacitance, keeping the plate tank tuned,
until the plate current is approximately 100
milliamperes. Increase the excitation level to
obtain 80 mil:iamperes of grid current. Finally,
adjust loading and tuning to obtain a resonant
plate current of 200 milliamperes, keeping the
grid current at 80 milliamperes. Varying the
excitation level and the plate loading will permit a 2.5 -to -1 ratio between plate and grid
current to be held. An exciter delivering less
621
than 15 watts may be used provided the loading is sufficiently reduced to maintain the
same ratio between plate and grid current.
Under voice operation, meter readings will be
one -half (or slightly less) than the steady state readings indicated above. If a tuned
cathode circuit is used, it is resonated for
maximum grid current on each band.
Figure 20
UNDER -CHASSIS VIEW OF 7094
GROUNDED -GRID AMPLIFIER
Power supply components are grouped at left
side of chassis. The 0.01 pfd. ceramic bypass
capacitors are grouped about the socket to
keep all r.f. leads short. RFC, is mounted
directly on the socket between two pins. Filament transformer T, is at the right, with the
rectifier filament transformer mounted to the
rear wall of the chassis. Millen ceramic sockets
are used for the high -voltage rectifier tubes.
www.americanradiohistory.com
622
29 -7
H.F. Power Amplifiers
The
"Tri- Bander"
Linear Amplifier
for 20 -15 -10
With the advent of the trap -tuned "tri bander" beam, many amateurs are concentrating their efforts on the 20, 15 and 10 meter
bands. In addition, low frequency operation
is often impractical for amateurs located on
small city lots and their activities must be confined to the higher frequencies, This linear
amplifier is designed for the amateur whose
principal interest lies in the 14 -30 Mc. spectrum. An amplifier built specially for this
range can be made smaller and more inexpensively than one that covers the complete
3.5 -30 Mc. range.
The unit described in this section is a one
kilowatt p.e.p. class ABt cathode driven
linear amplifier using two compact, ceramic
4CX300A tubes. A novel and easily built
chassis- cabinet enclosure is employed, together
with the inexpensive model of the Eimac air system socket. The amplifier is small enough
so that it may be placed on the operating table
next to the sideband exciter and receiver. Pro-
T
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D
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visions are made for voice operation, or for
operating the s.s.b. exciter without the amplifier. At 2000 volts plate potential, third
order distortion products are better than -30
decibels below maximum signal input.
Amplifier Circuit
A high perveance tube
such as the 4CX300A
cannot be used in a conventional class B
grounded grid circuit, as the element geometry
leads to high grid current and to destructive
values of grid dissipation. The distortion reduction characteristics of grounded grid circuitry, however, may be retained in an acceptable cathode driven circuit, wherein grid and
screen operating potentials are applied to the
tube. The schematic of this amplifier which
makes use of such a circuit is illustrated in
figure 22. Two 4CX300A tubes are employed,
with the driving signal applied to the cathode
circuit as is done in the common grounded
grid configuration. Grid and screen elements
are at r.f. ground, while normal Class AB'
grid bias and screen potentials are applied to
the tubes. Under these conditions, the power
gain of the 4CX300.A is quite high; approxi-
Figure 21
TRI -BANDER LINEAR
AMPLIFIER FOR
10 -15 -20 METER
SIDEBAND
kilowatt p.e.p.
linear amplifier is deThis
one
signed for those amateurs interested in the
higher
frequency DX
bands. Using two 4CX300A tubes, this compact
bandswitching
unit is
ideally suited for exciters
having a p.c. p. output of
about 30 watts. Panel
controls are (I. to r.):
Screen meter, plate meter,
plate tuning, plate loading. On the left is the
mode switch, Si; and on
the right is the band
switch, S2. Amplifier is
mounted on four rubber
"feet" so that cooling
air may be withdrawn
from under the cabinet.
Geared tuning
dials,
switch knobs, and plate
bandswitch are salvaged
from surplus "TU" tuning drawers from BC191;375 transmitter.
r
www.americanradiohistory.com
HANDBOOK
Tri -Bander Amplifier
E
623
AC.
R.P.
RY
INPUT
OUTPUT
RFC,
B+2 NV. 0
500
Hy
=
SIA
sew*1
0=
-BIAS
o
500
KV
T,0
1_
°RFC
RFC3
u
50
250
C,
52
C2
RFC1
2
ITT
RFC,
3
T
4
000
RFC.
,
ANT RELAYQ
B+ SCR.
-r
-_ll
_
T
RFCI
_
,OOK
RFC3
2K
X
115
AT EACH
F0rT
VA, i25t)
C
o,
RFCI
SOCKET
Sie
4
1
MA>
E
AU*.^
RFC,
B
CONTROL
o
TUNE
T
1rs
SWITCH
RFC,
6001
CONTROL
MS-1
O
B
CONTROL
Off
2
AUX.
NOTE
1
:
1.
CAPACITORS -C-ARE .0011/F., 400 V,
DISC CERAMIC.
2. ONE CAPACITOR "C" ON EACH GRID
AND SCREEN SOCKET TERMINAL (1.1-)
3. PI LS.
C
T
1.
3.-
4. TUNE
5. OPERATE
INDICATES PEEDTHRU CAPACITOR
-11I
9
CND.
Figure 22
SCHEMATIC, TRI -BANDER LINEAR AMPLIFIER
-50 III,fd., 3 kv.
155 -8 (50F30), 0.075"
-250 /IIId., 2 kv.
Johnson
spacing
Johnson
155 -6 (250F20), 0.045" spacing
L, -10 mete,. section: 31/2 turns,
3/16" copper tubing, wound
11/4" i.d. Adjust length to
resonate with C1 25% meshed.
15 -20 meter section: S turns,,
Vs" copper tubing, wound
21/4" i.d. 15 meter tap 3
turns from "cold" (output)
end
M,
-5C d.c. milliammeter. Recalibrated to -20 to +30 ma.
C1
C2
-0
M2-0 -S00
MS1 -SPST
switch"
d.c. milliammeter
lever -type
"Micro -
PC-Parasitic
choke. Two turns
diem., wound
1/2 -inch
about 47 ohm, 2 watt composition resistor
RFC1 -VHF choke. Ohmite Z -144
RFC2 -44 ph., S00 ma., Ohmite
«12,
Z -14
RFC3
-2.5
tional
mh,
R -300
RY1 -DPDT,
115
tenna relay.
2C-115VA
300
ma.
Na-
volt coil, anAdvance AM-
Si-Two
pole, 5 position progressively shorting switch. Two
Centralab « P -1 decks, with
P -121 Index Assembly
S2
Single -pole, 5 position ceramic switch from surplus "TU"
tuning unit, or Centralab
--
T,-6.3 volt at
6
amp. Stancor
Adjust primary resistor to deliver 6.0 volts at
tube sockets under load
Blower-35 cubic feet per minute. 6000 r.p.m., 115 volts
P -6456.
a.c. Ripley «8445 -E
Feedthrough capacitors -Each of
the eight control leads, plus
the two leads to the relay
coil pass through 0.001 Aid.
ceramic feedthrough capacitors. Centralab type FT -1000
Sockets: Eimac 5K -760 air socket.
Place one 0.001 Aid., 600
volt ceramic capacitor from
each screen
terminal to
ground
«2550
mately 30 watts p.e.p. drive being required
for full output.
The amplifier plate circuit is a simple three
band pi- network, designed for a circuit Q of
15. As the low frequency bands are not in-
cluded, only two small self- supporting air
wound coils are required. In addition, the size
of the pi- network loading capacitance is considerably smaller than a capacitor necessary for
all band operation.
www.americanradiohistory.com
624
H.F. Power Amplifiers
The amplifier is controlled by a two deck
progressively- shorting switch (S1) that remotely controls the auxiliary equipment and
provides the operator with a choice of "tune"
or "operate" modes. All control and low voltage power leads are suitably filtered by L -C
networks to suppress radiation of TVIproducing harmonics.
The "Tri- bander" linear amplifier construction is novel in that no regular chassis deck is
employed. The amplifier is built within an
enclosure made up of two aluminum chassis,
each measuring 10" x 14" x 3 ". One chassis
is inverted and serves as a pan within which
the components are mounted. The second
chassis is placed atop the first and serves as a
top shield cover. This chassis assembly is
hinged along the rear edge, and opens up much
in the manner of a suitcase. A single -piece
front panel made of aluminum is fixed to the
lower chassis. The front apron of the top
section is cut away to provide clearance for
the meters, switches and capacitors. When the
top section is closed, the cabinet is sealed by
a strip of finger stock that runs around the
inside edges of the lower chassis box. A length
of "piano -type" hinge fastens the rear edges
of the two chassis together, and the enclosure
halves are held in place by five panel bolts
which screw into nut plates riveted to the lip
of the lid, or top section.
An aluminum partition divides the interior
of the enclosure into two compartments
(figure 23). The smaller compartment contains the blower motor, filament transformer,
panel meters, auxiliary control relay, function
switch, and power lead filters. The larger compartment contains the two 4CX300A tubes,
the plate circuit pi- network components and
the antenna relay. The partition is shaped to
fit around the housing holding the tetrode
tube sockets. As the standard air system socket
with built -in screen bypass capacitor is both
expensive and bulky, the smaller phenolic
socket having no screen capacitor was used as
an inexpensive substitute. Two of these sockets
will mount atop an oscillator shield can taken
from a defunct surplus "Command" transmitter. The can makes an inexpensive and r.f.tight shield for the grid and cathode components, and is mounted directly to the bottom
chassis "pan." The pi- network capacitors and
bandswitch are panel mounted, and the re-
T H
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maining compartment area is taken up by the
plate coils, r.f. choke, and the plate blocking
capacitors. Antenna relay RY, is mounted
within a small aluminum shield box placed at
the back of the compartment.
Transmitter wiring is simple and straightforward. All connections in the meter compartment are made with unshielded wire. The
relay leads pass through the internal shield
partition via high frequency feedthrough
capacitors, and the exciter switching leads to
the contacts of the relay pass through short
lengths of RG -58/U coaxial line. The outer
braided conductor of the line is soldered to a
u.h.f. -type "hood" (Amphenol type 83 -1H) to
ensure r.f.-tight connections where the cables
enter and leave the amplifier compartment.
The three ceramic capacitors that make up
the plate blocking unit are mounted atop the
plate r.f. choke, and are fastened to the main
tuning capacitor by means of an aluminum
strap visible in figure 23.
Connection is made to the anode of each
tube by means of a 1/2 -inch wide copper strap
encircling the air cooler structure. Air is
drawn through 1/4 -inch holes in the bottom
pan by the blower, forced into the grid compartment, circulated upward through the tube
socket and cooling anode, and exhausted via
1/4-inch vent holes drilled in the top lid of
the enclosure. The blower motor goes on whenever the filaments of the tubes are lit.
Transmitter Control
Circuits and
S1 controls the
transmitter and auxiliary
Power Supply
equipment. All circuits
are off in the first position. In the second position, an auxiliary circuit is completed which can turn on the station
receiver or sideband exciter. The third position
turns on the amplifier tube filaments and
energizes the blower motor to cool the tubes.
Cut -off bias is applied to the tubes to eliminate
diode noise often noticed in standby operation.
The fourth position applies full plate voltage
and reduced screen voltage to the amplifier for
tuning operations, and the fifth switch position
applies full screen voltage. Cut -off bias is removed by the voice -actuated relay in the power
supply. Screen and plate currents are continually monitored by the two panel meters.
The screen meter is recalibrated to have an
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Switch
HANDBOOK
Tri -Bander Amplifier
625
Figure 23
INTERIOR VIEW OF LINEAR AMPLIFIER
The r.l. components are contained in the compartment to the right of the shield partition. Antenna relay RYA is placed in small aluminum box mounted to rear wall of
cabinet directly behind antenna loading capacitor. The two 4CX300A tube sockets are
mounted on top of aluminum shield can taken from oscillator coil section of surplus
"command" transmitter. Micro- switch on partition removes high voltage when cover is
opened. Midget relay adjacent to switch is added for auxiliary control circuits and is
not required. At extreme left rear cre feedthrough capacitors mounted on aluminum
plcte, with r.l. chokes beneath them. Filament transformer is in corner of compartment,
in back of mode selector switch. Pi- network components are at right, with three plate
blocking capacitors mounted to aluminum strip supported by plate tank capacitor.
elevated zero point and reads -20 to +30
milliamperes. Under certain conditions, negative screen current can flow and it is important
to monitor this sensitive indicator of amplifier
operation.
The power supply schematic is shown in
figure 24. The high voltage supply uses 3B28
"hash-free" gas rectifier tubes and provides
2000 volts d.c. at 500 ma. and regulated 360
volts at 30 milliamperes. "Jumpers" in the
base of the regulator tubes are wired in series
with the primary relay circuit so that the
supply cannot be energized unless the tubes
are in their sockets. A smaller half -wave semiconductor supply provides operating and cutoff bias for the amplifier. The bias relay may
be actuated by the voice circuit of the exciter
to drop the bias to the correct amount during
the time the voice circuit is energized.
www.americanradiohistory.com
THE RADIO
H.F. Power Amplifiers
626
T,
B+ 211V.
CH1
RY 2A
6-1-SCR. (360 V.)
(s3)
20
0.15
12
411V
311V
Roo
R105
GN D.
(a9)
VR105
n5V.'NJ
SR
T3
p
p
+ 100
100
ADJ.
BIAS
GRID
2W
1011
RESISTOR
°
20
I
4W
BIAS
(a4)
METER
4
2W
.
7
11
01,5
V.
WI
(ell
RY2B (a21
RY2
RY4
VOX
TO PR
- MANUAL
JUMPERS IN VR TUBES
7
3
7
3
7
CONTACTS
at S00 ma. Chicago R -65
lamp and receptacle
RY2 -DPST, 11S volt coil. Potter -Brumfield
MRSA, 115 volt a.c.
RY3 -SPST, 115 volt coil, 20 amp. contact.
Potter -Brumfield PR3AY, 115 volt, a.c.
RY4 - -DPDT, 115 volt coil. Potter -Brumfield
MR11A, 115 volt a.c.
SR- Selenium rectifier, S00 ma. Sarkes- Tarzian
h.
(a7)
O OPERATE CONTROL
Figure 24
SCHEMATIC, LINEAR AMPLIFIER POWER SUPPLY
M -500
-2.5 volts at 10 a.,
Chicago F -210H
T2- 2900 -2300 volts each
10
kv.
insulation.
side of c.t. at 500
ma. 115 -230 volt primary. Chicago P -2126
v., 50 ma. Stancor PA -8421
Extra contact set of RY4 is placed in series
with antenna relay control lead (17 2) and
-125
RY2R
CONTROL
-115 volt pilot
T1
T3
(a6)
V.*2
X
(.6)
CONTROL RELAY
P3
I
15
-A U
TUNE' CONTROL
3
P,,
ECE
0-O
C11TVER
SWITCH 6--ce-
TO VOX
CH1-6
R
1.
ANT. RELAY
contacts to actuate antenna relay
R Y,
(figure 22) by VOX circuit.
The only initial adjustment is
to set the operating bias level
by means of the potentiometer.
Initially, the arm should be set
at the high potential end of the potentiometer
to apply full bias to the tubes. The filaments
and blower are turned on, and the high voltage and bias supply energized. Using a voltmeter, the potentiometer should be set to
provide about -60 volts on the arm. The
voice relay is energized dropping the cut-off
Transmitter
Adjustment
and Tuning
bias out and the potentiometer is carefully
reset to provide a static plate current of 200
ma. as read on the meter. Indicated screen current (bleeder current) should be about 22
ma. When the voice relay drops out, the plate
current should fall to zero.
The amplifier is now fed a small exciting
signal (single tone) and tuned and loaded for
a maximum plate current of 500 milliamperes.
Screen current should now be approximately
30 ma. (This is a total of screen and bleeder
current.) The output coupling is now increased
slightly so that r.f. output (as read on an r.f.
ammeter, or output voltmeter) drops about
2 percent. Maximum linearity is obtained when
the amplifier is slightly overcoupled. Under
voice conditions, plate current peaks should
reach approximately 250 ma., as read on the
meter. No grid current should be read on a
0 -1 d.c. milliammeter placed across the grid
current terminals in the power supply. Any
flicker of grid current indicates the amplifier
is being overdriven, with a consequent severe
www.americanradiohistory.com
HANDBOOK
813 Linear Amplifier
increase in distortion. Under voice conditions,
indicated screen current will be relatively constant, as actual current drawn by the screen
of the tubes will be less than + or
10 ma.,
and this small value is swamped out by the
bleeder current, which is constant at 22 ma.
Low values of screen meter current (indicating
that the tubes are drawing negative current)
indicates excessive loading; high values of
screen current indicate insufficient plate circuit
-
loading.
Never apply excitation to this (or any
other) grounded grid amplifier without all
operating potentials applied to the tubes.
Figure 25
THE 813 GROUNDED -GRID LINEAR
AMPLIFIER
Two 813's are used in this simple and effective linear amplifier. Built on a 101'2 -inch
rock panel, the amplifier may be placed in a
metal cabinet for desktop operation. Capable
of operation on all amateur bands between
80 and 10 meters, this unit may be driven by
the popular 75 to 100 watt sideband exciters.
Panel controls are (I. to r.): bandswitch, plate
tuning (top) and antenna loading (bottom),
meter switch (top) and bias control (bottom).
Front bushing of linkage shalt for switch S2
passes through panel between tuning and
loading controls and is camouflaged with
small knob.
29 -8
627
An 813 Grounded Grid Linear Amplifier
The popular amateur s.s.b. transmitters in
the 75- to 100 -watt power class provide a
ready -made exciter when the time comes to
add a more powerful final amplifier to the
amateur station. Because tetrodes have low
power drive requirements, a power dissipating
device must be employed when these tubes are
driven from a 100 -watt class transmitter. A
suitable dissipation device is usually fragile,
expensive, and difficult to construct. In addition, the tetrode tube requires bias and screen
power supplies which are bulky and expensive.
A grounded grid amplifier circuit provides
a satisfactory solution to these problems as no
power dissipating device is required, and
screen and bias supplies may be eliminated.
Certain tetrodes and pentodes operate well as
zero -bias, grounded grid triodes, and the 813
is one of these. This tube operates efficiently
in class B grounded grid service at plate poten-
www.americanradiohistory.com
628
THE RADIO
H.F. Power Amplifiers
LI
EA.
813
.001
813
57/V
7L15_2
500
L.
=
1.2Kv
Tool
C2
KK
V
RFC2
II
FIL. CET.
1500
0
50Ó
t
CAPAC ITOR
RFC
50-760
RFC4
r.
OUT.
C1
10 KV
.ó1
R
13+ 14.V.
I
325
CAI
ADJ. B/AS
R,
ALTERNATIVE TUNED
CATHODE CIRCUIT
.01 .01
TS2
1.
52
ITS,
O
Io
ro
2
VOX
RELAY
3
NOTES
t 40M POSITIONS. OPEN /1120,
POSITIONS OF BANOSW/rCN SI.
CLOSED IN 80
154
10 M.
2. ALL 01 CAPACITORS ARE BOO V CERAMIC UNITS.
3. JUMPER TERMINALS / B 2 ON TERMINAL STRIP
ro REMOVE BLOCK /NG B /AS.
4
115 V.1,
CND
Figure 26
SCHEMATIC, 813 LINEAR AMPLIFIER
B- Tube -cooling
motor and fan. Shaded pole
induction motor, 2400 r.p.m., with 4 -blade
fan, 21/2" diom. Allied Radio Co., Chicago.
Part 5 72P715
C1- Two- section variable capacitor. Front section (added for 40 -80 meters): 28 -160 ppfd.
Rear section: 7-50 ppfd. 0.125" spacing.
Barker & Williamson. A conventional split stator capacitor may be substituted. Johnson 1;l54 -3 (100E045) is recommended. Install the switch between the stators, on the
studs supporting the stator plates at the
middle of the capacitor. Change length of
linkage to fit new layout.
C2 -1500 ppfd., 0.03" spacing. Barker & Williamson
51241. A four section, b.c. -type
variable capacitor (J. W. Miller ,V.2l04) with
sections in parallel may be substituted.
C3 -1260 µµId. Three section b.c. -type capacitor (J. W. Miller 52113) with sections in
parallel
C4 -325 µµId., 0.024" spacing. Hammarlund
MC -325M
L, -10.5 µh. transmitting inductance. Barker
& Williamson 850A. Air -Dux 5195 -2 coil
may be substituted. This coil should be
trimmed and topped to resonate as follows:
80 meters, 210 ppfd.; 40 meters, 105 ppfd.;
20 meters, S2 ppfd.; 15 meters, 30 µpfd.;
10 meters, 30 ppfd. Above capacities include output capacitance of tubes
-10
L3
meter section: 0.44 ph. S turns 512
diam., 1" long, space -wound S turns
per inch. Tapped section: 4.2 ph. 17 turns
5 16 tinned, 11/2" diom., 21/4" long, space wound 8 turns per inch. Tapped 2 (21 Mc.),
4 (14 Mc.), and 10 (7 Mc.) turns from 10
meter end of coil. B&W 53018 miniductor
e., 1"
-0
-1 d.c. milliammeter
RFC, -0.5 mh., 300 ma.
M,
R -300
RFC2
-15
choke.
National
ampere filament choke. B&W type
FC -15
RFC3
-200
ph.
choke.
National R -175A
RFC4,3-1 mh., 300
type
800, or
choke.
National
B&W
ma.
R -300
-Part
of L,. An Ohmite type 111 -5, S
position ceramic switch may be used with
Air -Dux coil. Switch should be mounted on
an insulated brocket and driven with an
insulated coupling
S2- Special switch. See text for details
S3- Single -pole, S position ceramic. Centralab
52500
SR -130 volt, 75 ma., replacement -type selenium rectifier
S,
-10
volt, 10 amperes. Thordarson 21F19
volt, 50 ma. Stancor PA -8421
711,2-Insulated terminal strips. Cinch -Jones
T,
T2
-115
Knobs -B&W 5901 (11/2" diam., 3 req.) B&W
»903 (11/16" diam.,
www.americanradiohistory.com
3
req.)
HANDBOOK
813 Linear Amplifier
629
Figure 27
LEFT REAR VIEW OF AMPLIFIER
A I á -inch thick sheet of aluminum 13 inches by 17 inches in size forms the main chassis and is
fastened to the panel with chassis support brackets. Connection between plate r.f. choke, blocking
capacitors, plate tuning capacitor and plate coil are made with copper strap. Plate leads from
tubes to strap are made with =10 flexible braided wire. Coaxial r.f. input receptacle is next to
11S -volt line cord, and antenna receptacle is mounted on angle bracket at end of sub -chassis.
Switch S, is at rear of bandswitching inductor.
bals up to 3000 volts. Two 813's in parallel
at 2500 volts will provide a p.e.p. input of
1500 watts (750 watts, single tone) provided
cooling air is circulated about the tubes. At
3000 volts, a p.e.p. input of 2000 watts (1000
watts, single tone) may be run but the plate
dissipation of the tubes exceeds the recommended maximum figure. If plenty of cooling
air is used, this does not seem to shorten tube
life. Under these two operating conditions,
third order distortion products are better than
-30 decibels below maximum power level.
Amplifier Circuit
The circuit of this linear
amplifier is shown in
figure 26. The basic amplifier employs an untuned cathode input circuit for simplicity and
low cost, although an alternative tuned input
configuration is shown. Improved intermodulation distortion suppression and less driving
power can be gained with the use of the tuned
circuit.
The screen and beam -forming plates of the
813's are grounded directly at the socket. The
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630
THE RADIO
H.F. Power Amplifiers
grids are bypassed to ground and receive a
small amount of negative bias from the built in bias supply. The exact bias level may be
set by the potentiometer. In addition, when
the connection between terminals 1 and 2 on
the terminal strip is broken, the tubes are
biased to cut -off to eliminate troublesome diode
standby noise. When these terminals are
shorted by the contacts of the voice relay, the
bias is reduced to the operating value determined by the setting of the potentiometer.
Separate metering of current in the grid and
plate circuits is accomplished by switching a
single meter (M) across shunt resistors. The
0 -1 d.c. milliammeter is converted into a low range voltmeter by the addition of the 1.2K
series multiplier resistor, and the voltage drop
across grid and plate shunt resistors is measured. In the grid position, the meter reads
0 -100 ma., and in the plate position it reads
0 -500 ma.
A pi- network plate tank circuit is employed.
Optimum plate load impedance for this circuit is about 5000 ohms, and the Q should
be held to a figure of 15 or better. These
requirements may be met with the specified
components, or with less expensive substitutes,
as outlined in the parts list.
High voltage is applied to the parallel -connected 813's through the plate r.f. choke. Three
blocking capacitors in parallel keep high voltage from reaching the pi- network plate tank
circuit. A tapped coil and two section tuning
capacitor provide nearly optimum L/C ratio
on all amateur bands from 80 to 10 meters.
Only one section of the tuning capacitor is in
the circuit on the 10, 15 and 20 meter bands
when the automatic switch S2 is open. Both
capacitor sections are in parallel on 40 and 80
meters where greater maximum tuning capacitance is required, S2 being closed by a mechanical linkage from the main bandswitch, S,.
A large variable pi- network output capacitor
(1500 µµfd.) eliminates the need for several
fixed capacitors and a tap switch to add them
to the circuit as needed. The output circuit
will match load impedances in the range of
50 to 75 ohms having an s.w.r. of 2/1 or less.
Figure 28
RIGHT REAR VIEW
OF AMPLIFIER
Main tuning
are mounted
end -brackets
!Vs -inch sheet
capacitors
on vertical
made of
aluminum.
The copper nngle brackets on the plate capacitor plus U- shaped bracket
on switch linkage form
foreground, mounted on sub -chassis are
the filament transformer,
bias supply filter capacitor, high voltage terminal, and plate r.f.
choke. Bottom chassis
plate is drilled beneath
fan to permit cooling air
to be drawn into sub Sy. In
chassis area.
www.americanradiohistory.com
HANDBOOK
813 Linear Amplifier
631
TOP VIEW
REAR SUPPORT
PLATE IOR CI
AND
C2-AY 7Y}
} f
ALUMINUM
Y
SO
PANEL
FRONT PLATE
!BASS STRIP
E-
LUCITE
2
LONG
+LONG
LI
ON
U
-CLIP
FROM
FORMED FROM
it2Y=r
!BASS
1
SPRING
LONG
FRONT SUPPORT
FOR
C2
CI
2
If 7
sRAS3 STRIP
PANEL
LONG
POSITION OF LINRAGC
IN 14,21 ANC 211-MC
POSITIONS OF
LI"
Alt
m;p=A=1R>=::
Vir
POSITION
LINKAGE
AND
OF
1
MC
LIST
O
IM
3
].]\
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POSITIONS
FRONT VIEW
Figure 29
DETAIL DRAWING OF
SWITCH S, LINKAGE
Three ! /e" x /2" brass strips, soldered to brass shaft couplings
make up the linkage arms. Plastic
arm supports U -clip which closes
circuit between copper angle brackets mounted on main tuning capacitor in 80 and 40 meter positions
of bandswitch.
Amplifier
Amplifier construction is quite
simple due to the utilization of
standard, readily available components. The main chassis is a 14" x 17" x
1,á -inch thick sheet of aluminum fastened with
its bottom surface !48-inch above the lower
edge of a 101/2" x 19" aluminum relay rack
panel. Only the pi-network components, meter,
and meter switch are mounted to the main
chassis, the remaining components being assembled on the 6" x 11" x 21/2" aluminum sub chassis. The photographs and drawings illustrate the placement of the major components.
The end plates of the tuning capacitors are
-inch aluminum brackets seven
fastened to
inches high and four inches wide (figure 30).
The shaft on which the linkage for switch S2
is supported also runs between these brackets.
Construction
/
The parts of this linkage, and assembly details
are shown in figure 29. A U- shaped clip, made
from spring brass or phosphor bronze, completes the connection between copper angle
brackets fastened to the two stator sections on
the main tuning capacitor when the bandswitch
is in the 80 and 40 meter positions. The short,
rotary arm on the bandswitch is adjusted so
that it engages the forked arm, as shown in
solid lines in the sketch when the bandswitch
is in the 40 meter position. Both arms should
then move up so that the forked arm is in the
position indicated by the dotted lines when
the bandswitch is in the 20 meter position.
The rest of the plate circuit wiring is done
with silver plated ', -inch copper strap. The
strap is ordinary flexible copper "flashing" cut
into strips and silver plated by a local utensil
replating company.
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632
H.F. Power Amplifiers
PANEL LAYOUT
T H
BRACKET FOR
AMPLIFIER
A D
I
O
,
CI 1.C2
Figure 30
PANEL LAYOUT FOR
AMPLIFIER
PLATE
CITUNING
I
R
SUPPORT
813 GROUNDED GRID
2
E
DIA KNOBS
TUNT
r,
HOLE DIA
22
¡LARGER
THAN METER
--J-L-1
GRID
ALOA ING
MET
-1
ri
PLATE
A
I
I(
t
SW
S3
RI BIAS
CASE
I
2
LaW)
L`-
linkage for capacitor
switch pivots on shaft located between main tuning
capacitors. Drill 3/4-inch holes
for this shaft, and the shafts
of the capacitors, plus the
meter s w itch. Aluminum
is
chassis -deck
positioned
I9 -inch above bottom edge
of panel.
The
2Ii3
S
4
Sub -chassis assembly and wiring is shown
in figure 31. The ceramic sockets for the 813
tubes are sub -mounted on metal pillars to
bring the top of the socket shell level with the
under side of the top of the chassis. Under chassis wiring, with the exception of the #12
filament leads is run with #18 insulated wire.
The filament choke and bias transformer are
mounted on opposite walls of the chassis. A
small, 115 volt blower motor and fan draws
air up through 1/4-inch holes drilled in the
bottom chassis plate and exhausts the air
through the holes cut in the sub -chassis for
the 813 tubes.
Socket pins 3 and 5 are connected together
and grounded to each of the two adjacent
socket bolts. A jumper runs between the #4
pins, each of which are bypassed to ground by
a .001 µfd. ceramic disc capacitor. Each capacitor must be a 1.2 KV type in order to carry
the r.f. charging current existing in the grid
circuit. In addition, a small 50 gefd, ceramic
capacitor is connected between pins 1 and 4
of the tube socket nearest the filament choke.
This capacitor stabilizes the amplifier in the
28 Mc. region.
The 10 volt filament transformer for the
813's is placed above the chassis, as are the
plate r.f. chokes and bypass capacitors. The bias
filter capacitor is a can -type unit which mounts
adjacent to the filament transformer. Various
meter leads are brought out of the chassis via
a terminal strip mounted on the side opposite
the power cable and coaxial input receptacle.
4
In a TV fringe area, it may be necessary to
completely shield the amplifier with perforated
aluminum sheet. Amplifier harmonic content
is low, and complete shielding is not necessary
in an area of strong TV signals.
Testing and
Operating the
Once construction is finished,
check the filament and bias
Amplifier
circuits before connecting the
high voltage supply to the
amplifier. A power supply with provision for
reducing the output to about one -half of maximum voltage is recommended, especially if
the operating voltage is 2500 or higher. Connect a dummy load or antenna to the output
receptacle.
Caution: Never apply full excitation to this
or any other grounded grid amplifier without
the plate circuit tuned to resonance, and plate
voltage on the stage. Damage to the amplifier
tubes may result if this rule is violated.
Tune-up for sideband operation consists of
applying full plate voltage and (with terminals
1 and 2 on the power strip shorted) setting
the bias potentiometer for 55 milliamperes of
resting plate current with the meter switch
set in the "plate" position. Only a few volts
of bias are required, and the potentiometer arm
will fall very near one end of the swing. Set
the bandswitch to the frequency of the exciter
and apply a small amount of driving power by
injecting carrier in the s.s.b. exciter. Place the
loading capacitor at full capacitance, and adjust the plate tank capacitor for resonance
(minimum plate current). Apply more drive
www.americanradiohistory.com
HANDBOOK
813 Linear Amplifier
Figure 31
TOP AND BOTTOM V EWS OF SUB- CHASSIS
Filament transformer and filter capacitor are placed at left edge of chassis. 813 socket holes are
frcm opposite end of chassis. Small plate choke is
2 -9, 16- inches in diameter, placed 214- inches
supported on bypass capacitor terminals. Bias transformer and filament choke are mounted to
underside of chassis, as
is
blower fan.
www.americanradiohistory.com
633
634
THE RADIO
H.F. Power Amplifiers
to obtain about 75 ma. grid current and decrease the loading capacitor until resonant
plate current rises to about 200 ma. Finally,
increase the drive and increase the loading
until plate current reaches 400 ma. (300 ma.
at a plate potential of 3000 volts). Grid current should be approximately 100 ma. Slightly
overcouple the antenna circuit until the output
(as measured on an r.f. ammeter) drops about
2 percent. This will be the condition of maximum linearity. Now, switch the exciter to
s.s.b. With speech, the plate current of the
linear amplifier should kick up to about 135
to 150 ma.; while with a steady whistle the
plate current should reach nearly 400 ma.
Tune -up for c.w. operation is similar, except
that the bias potentiometer is adjusted for zero
(cut -off) resting plate current. With full plate
voltage (2500) , the resonant plate current
should be about 375 ma., with 100 ma. of
grid current. At a plate potential of 3000, the
plate current should be reduced to 300 ma.
29 -9
The KW -2. An
Economy Grounded Grid Linear Amplifier
The KW -2 sideband amplifier is designed
for use with 4 -400A, 4 -250A or 4 -125A tubes,
and will operate on the 80, 40, 20, 15 and
10 meter amateur bands. A pi- network output
circuit is used, capable of matching 52 -ohm
or 75 -ohm coaxial antenna circuits. Maximum
power input is 2 kilowatts (p.e.p.) or 1
kilowatt, c.w. The amplifier may be driven by
any of the popular s.s.b. exciters having 70 to
100 watts output.
Full input may be achieved with the use
of 4 -400A tubes, but the unit may be run at
reduced power rating with 4 -250A or 4 -125A
tubes. No circuit alterations are necessary when
tube types are changed.
The amplifier employs a passive (untuned)
input circuit, and an adjustable pi- network
output circuit. Air tuning capacitors are used
in the network in the interest of economy and
Figure 32
REAR VIEW OF
AMPLIFIER PLATE
CIRCUIT
Sub- chassis has been removed to show ventilation holes in chassisdeck. Plate bypass capacitors are supported by
t/ -inch copper strap
leads.
www.americanradiohistory.com
HANDBOOK
KW -2 Amplifier
with no sacrifice in performance. The complete
amplifier is housed in a TVI- suppressed perforated metal cabinet measuring 171/4" x 12"
x 121/i"
small enough to be placed on the
operating table next to your receiver.
Amplifier Circuit. The schematic of the amplifier is shown in figure 34, Two tetrode
tubes are operated in parallel, cathode driven,
with grid and screen elements grounded. The
sideband exciting signal is applied to the filament circuit of the tubes, which is isolated
from ground by an r.f. choke. The resistance
of the windings of the choke must be limited
to .01 ohms or less, as filament current is 30
amperes for two 4 -250A or 4 -400A tubes.
Neutralization is not required because of the
excellent circuit isolation afforded by the
grounded elements of the tubes.
The Input Circuit. The input signal is fed in
a balanced manner to the filament circuit of
the two tubes. Ceramic capacitors are placed
between the filament pins of each tube socket,
and excitation is applied to each tube through
two 1250 volt, mica capacitors. The latter are
employed because of the relatively high value
-
635
of excitation current which may cause capacitor heating if ceramic units are employed
at this point.
The filament circuit is wired with #10
stranded insulated wire to hold voltage drop
to a minimum. The leads from the choke to
the filament transformer are run in shielded
loom which is grounded to the chassis at each
end of the wire. The use of shielded leads for
all low voltage d.c. and a.c. power wiring does
much to reduce TVI -producing harmonics.
Figure 33
KW
-2
LINEAR AMPLIFIER
THE
This two kilowatt p.e.p. amplifier uses two
tubes in a grounded -grid circuit. Other
tetrodes, such as the 4 -125A and 4 -250A
may be used without modification to the unit.
At full output, distortion products are better
than -30 decibels below peak power level.
Panel components are (I. to r.): Plate current
meter (top) and output meter (bottom), meter
switch and pilot lamp, plate tuning, band switch, and plate loading. At lower right is a
tuning chart for the various bands.
Chassis is bolted directly to the front panel,
allowing about
-inch clearance along bottom
edge to permit edge of shield cage to pass
between chassis and panel lip.
4 -400A
/
s
v
www.americanradiohistory.com
636
THE RADIO
H.F. Power Amplifiers
4 -400A
4 -400A
ANT
exc
N OTE:
RI +METER RES /STANCE
.
100.11
C2
C
Figure 34
SCHEMATIC, KW -2 AMPLIFIER
-0.001 pfd., 600 volt disc ceramic
C,,C2,C3 -0.1 pfd., 600 -volt coaxial capacitor.
Sprague "Hypass" z80P3
C4 -150 µpfd., 4500 volt. Johnson =1501345
(153 -8)
Cs -SO µµId., surplus vacuum capacitor (see
C
The Grid Circuit. The grid circuit of this amplifier is simplicity itself. Screen terminals of
both sockets are grounded to the chassis of the
amplifier. The best and easiest way to accomplish this is to bend the terminal lead of the
socket down so that it touches the chassis.
Chassis and lead are then drilled simultaneously for a 4 -40 machine screw. Low inductance ground paths are necessary for the high
order of stability required in grounded grid
service.
It is helpful to monitor the control grid
current for tuning purposes, and also to hold
the maximum current within the limits given
in the data chart. Maximum grid current for
the 4 -400A is 100 milliamperes. Under normal voice conditions this will approximate a
peak meter reading of 50 milliamperes.
Grid current can be observed by grounding
the control grid of each tube through a 1 -ohm
composition resistor, bypassed by a .01 pfd.
disc capacitor. The voltage drop across the
text)
Ce
-1000
text)
µµId., 1250 -volt mica capacitor (see
C7-1500 f.µfd. Barker
& Williamson »51241
2104
pi- network coil. Air -Dux #195 2S (silver plated). Modify as follows: Strap
coil: 3 turns, 13/4" diameter. Wire coil: Remove turns from free end, leaving 111/2
turns, counting from junction with tubing
coil. Tap placements: 10 meters, 13/4 turns
from junction of tubing coil and strop coil.
IS meters, 31/4, as above. 20 meters, 11/2
turns of wire coil, counting from junction
with tubing coil. 40 meters, 53/4, as above.
80 meters, complete coil in use
RFC1 -30- ampere filament choke. B&W zFC-
or 4 -gong b.c. capacitor. Miller
L,-Kilowatt
30
RFC2- Kilowatt r.f. choke. Raypar, or B&W
2800
RFC3- v.h.f. choke. Ohmite z I -50
T1-5 volts at 30 amperes. Stancor P -6468
PC -31/2 turns z 12e, r/e" diam., 2" long.
Wound around three 220 -ohm, 2 -watt composition resistors connected in parallel
M1-0 -1000 ma. Triplett
M2-0 -1 ma. Triplett
X1- Diode,
type
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11434
HANDBOOK
KW -2 Amplifier
resistor is measured by a simple voltmeter
calibrated to read full scale when 100 milliamperes of grid current are flowing through
the resistor. A double throw switch will permit
monitoring grid current of either tube. With
incorrect antenna loading, it is possible to
exceed maximum grid current rating with
some of the larger size s.s.b. exciters. No circuit instability is introduced by this metering
technique.
The Plate Circuit. Power is applied to the
plate circuit via a heavy duty r.f. choke bypassed at the "cold" end by a 500 µµfd., 10
kv. "TV -type" ceramic capacitor. In addition,
a v.h.f. choke and capacitor are used to suppress high frequency harmonics that might
pass down the plate lead and be radiated
through the power supply wiring. Two .001
cfd., 5 kv. ceramic capacitors in parallel are
used for the high voltage plate blocking
capacitor, and are mounted atop the plate
choke.
The
pi- network
coil
is
an
637
Air -Dux
#195 -2S inductance, designed for service at
a
kilowatt level, and silver plated for minimum
circuit loss. Use of the cheaper model having
tinned wire is not recommended for continuous service at maximum power. The band
switch is a Radio Switch Corp. #88 high
voltage, ceramic switch.
Figure 35
REAR VIEW OF AMPLIFIER
The tube sockets are placed at the right end
of the chassis, with plate r.f. choke centered
between the tubes. The two plate coupling
capacitors are mounted to top terminal of the
choke by means of a brass strap. A "TVtype" 500 0, fd. capacitor is placed at the
foot of the choke. The two panel meters are
mounted orte above the other. An aluminum
shield plate is placed around the rear of the
meters to protect them from the strong r.f.
field of the tubes. Meter terminals are bypassed, and the meter lecds are run in shielded
braid. Power, control terminals, fuse and
coaxial receptacles are mounted on rear apron
of chassis.
)
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638
THE RADIO
H.F. Power Amplifiers
SHAFT
OF
SWITCH S
CERAMIC PILLAR
I
,TOCS
Ir
SHAFT OF SI
SWITCH
ARM
to
f0 -32
TOP VIEW
4
`MAMI
BRASS BOLT
AND NUT
OF
/
TEFLON. PHENOLIC
OTHER INSULATING
TWO CONTACTS
SPRING BRASS.
,ALUMINUM BRACKET, BOLTED TO
OR
CORNER
MATERIAL
OF
C4 FRAME
SWITCH, SHOWN IN CLOSED POSITION
TOP VIEW
MODIFIED INSULATED
COUPLING.
SWITCH ARM
Figure 36
AUXILIARY PADDING SWITCH, PART
OF BANDSWITCH
Construction of padding capacitor switch made from parts of on insulated, flexible shaft coupler. Contacts are mode from 1/2 -inch wide
strip of spring brass mounted on small ceramic insulators attached to
main tuning capacitor. Contacts are shorted in 80 meter position of
bandswitch.
A circuit Q of 15 was chosen to permit a
reasonable value of capacitance to be used at
80 meters. In this case, a 150 µµfd. variable
air capacitor is employed for operation above
80 meters, and an additional 50 µµfd. parallel
capacitance is switched in the circuit for 80
meter operation. The 50 µµfd. padding capacitor is the small vacuum capacitor found in the
"Command" set antenna relay boxes. These
capacitors seem to be plentiful and inexpensive. A satisfactory substitute would be a
50 µµfd. 5 kv. mica capacitor, also available
on the surplus market.
The pi- network output capacitor is a 1500
µµfd. unit. It is sufficiently large to permit
operation at 80 meters into reasonable antenna loads. For operation into very low impedance antenna systems that are common on
this band, the loading capacitor should be
paralleled with a 1000 µµfd., 1250 volt mica
capacitor. This capacitor may be connected to
the unused 80 meter position of the band switch.
The Metering Circuits. It is always handy to
have an output meter on any linear amplifier.
A simple r.f. voltmeter can be made up of a
germanium diode and a 0 -1 d.c. milliammeter.
The scale range is arbitrary, and may be set
to any convenient value by adjusting the po-
tentiometer mounted on the rear apron of the
chassis. Once adjusted to provide a convenient
reading at maximum output level of the amplifier, the control is left alone. Under proper
operating conditions, maximum output meter
reading will concur with resonant plate
current dip.
It is dangerous practice to place the plate
current meter in the B -plus lead to the amplifier unless the meter is insulated from
ground, and is placed behind a protective
panel so that the operator cannot accidentally
touch it. If the meter is placed in the cathode
return the meter will read the cathode current
which is a combination of plate, screen and
grid current. This is poor practice, as the
reading is confusing and does not indicate the
true plate current of the stage. A better idea
is to place the meter in the B -minus lead between the amplifier chassis ground and the
power supply. The negative of the power
supply thus has to be "ungrounded," or the
meter will not read properly (figure 37) . A
protective resistor is placed across the meter
to ensure that the negative side of the power
supply remains close to ground potential.
Make sure that the negative lead between the
power supply and the amplifier is connected
at all times.
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HANDBOOK
KW -2 Amplifier
PLATE SWITCH OR
VOX RELAY
639
866'5
OR
3824.5
CONTROL
RELAY
T,
4
CHI
843000
V.
EA.
3011
50W
PRIMARY
CONTROL
SWITCH
100
20 W
CHASSIS
GROUND
15
IIS
0
ti
Figure 37
SCHEMATIC, POWER SUPPLY FOR LINEAR
AMPLIFIER
-6 H, 500 ma. Chicago R -65
T,-3450 -2850 volts each side of center
CH,
72
The Cooling System. It is necessary to provide
a current of cool air about the base seals and
plate seal of the 4 -250A and 4 -400A tubes.
If small blowers are mounted beneath each
tube socket it is possible to dispense with the
special air sockets and chimneys, and use the
inexpensive "garden variety" of socket. A
Barber Coleman type DYAB motor and impeller is mounted in a vertical position centered on the socket, and about an inch below
it. Cooling air is forced up through the socket
and around the envelope of the tube. The
perforated metal enclosure provides maximum
ventilation, yet effectively "bottles up" the
r.f. field about the amplifier. In order to permit air to be drawn into the bottom of the
amplifier chassis, small rubber "feet" are
placed at each corner of the amplifier cabinet,
raising it about 1/2 -inch above the surface
upon which it sits.
The amplifier is built upon an
aluminum chassis measuring
13" x 17" x 3 ". Input circuit
components, power circuits, and the blower
motors are mounted below the chassis, and
the plate circuit components are mounted
above the deck. Placement of parts is not
critical, except that the leads beween the bandAmplifier
Construction
tap, 500 ma. 115 -230
volt primary. Chicago P -3025
-2.5 volts, 10 a. 9 kv. insulation. Chicago
FH -210H
switch and the plate coil must be short, heavy
and direct. One -half inch, silver plated copper
strap is used. The straps are bolted to the
bandswitch with 4 -40 nuts and bolts. Each
lead is tinned and wrapped around the proper
coil turn and soldered in place with a large
iron. The operation should be done quickly
to prevent softening of the insulating coil
material. Low resistance joints are imperative
at this point of the circuit. To play safe, you
can submerge the coil in a can of water, with
just the top of the turns showing above the
surface. This will prevent the body of the coil
from overheating during the soldering process. It is also helpful to depress a turn on
each side of the tap in order to provide
sufficient clearance for the soldering iron. This
may be done by placing the blade of a screw
driver on the wire, and hitting it with a
smart tap.
The coil assembly is supported on four
ceramic pillars, and placed immediately behind the band change switch, which is
mounted on a sturdy aluminum bracket. The
coil is positioned so that the taps come off
on the side nearest the switch.
A set of auxiliary contacts are required to
switch the padding capacitor into the circuit
when the bandswitch is thrown to the 80
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640
THE RADIO
H.F. Power Amplifiers
r
t
urre
Figure 38
UNDER -CHASSIS VIEW OF AMPLIFIER
The filament transformer is mounted to the side apron, with the filament choke placed between
the transformer and the tube sockets. The two blower motors are attached to an aluminum strip
that holds them in position under the tube sockets, on a level with the bottom edge of the chassis.
This strip is bolted to the chassis flange with flat -head bolts. The bolts holding the blowers pass
through rubber grommets mounted on the strip to deaden blower noise. All low- voltage power
leads run through shield braid which is grounded to the chassis by means of aluminum clamps
mode from scrap material. B -plus lead is a section of RG -8 /U coaxial cable. Diode voltmeter components are mounted to a phenolic board attached to the side apron at right.
meter position. A simple switch may be made
up from the metal portions of an insulated
coupling and a block of insulating material,
such as teflon, lucite, or micarta (figure 36).
The insulated disc of the coupling is removed,
and an oval of insulating material is substituted. This assembly is placed on the shaft
of the bandswitch. A set of spring contacts
are mounted on small stand -off insulators
attached to the side of the tuning capacitor
and positioned so that the oval rotates between the contacts as the switch is turned. A
hole is drilled in the oval, and a flat -head
8 -32 brass machine screw is passed through
it. A nut is run onto the screw, and screw
end and nut head are filed flat. When the
www.americanradiohistory.com
HANDBOOK
KW -2
Figure
Amplifier
641
39
PLATE TANK CIRCUIT ASSEMBLY
The plate bandswitch is supported on a l's -inch thick aluminum bracket. The 80 meter padding
capacitor is mounted on the front of the bracket. Silver -plated copper strap is used to make
connections between the switch and the coil. Switch connections are made with 4 -40 hardware, and
then soldered securely. Auxiliary padding capacitor switch may be seen on shaft of bandswitch,
directly in front of bracket. Plate switch is made by Radio Switch Corp., Marlboro, N.I.
1
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642
H.F. Power Amplifiers
T H
E
R
A
D
I
O
Figure 40
OPERATING CHARACTERISTICS, GROUNDED -GRID CONFIGURATION
4 -125A
D.c. Plate Voltage
Zero -Signal Plate Current
Single -Tone Plate Current
Single -Tone Screen Current
Single -Tone Grid Current
Single -Tone Driving Power
Driving Impedance
Load Impedance
Plate Input Power
Plate Output Power
2000
2500
10
15
3000
20
105
30
110
30
55
115
30
55
55
volts
ma.
ma.
ma.
ma.
16
16
16
watts
340
10,500
210
340
3,500
275
340
15,700
345
240
ohms
ohms
1
190
145
watts
watts
4 -400A
(ratings apply to 4 -250A, within plate dissipation rating of
2500
3000
65
270
55
100
39
150
70
330
55
100
40
140
4500
675
435
5000
990
600
3000
4000
5000
100
120
150
700
675
540
105
170
130
104
80
150
105
106
55
115
2450
2100
3450
2700
5550
2700
1475
1870
1900
Zero -Signal Plate Current
Single -Tone Plate Current
Single -Tone Screen Current
Single -Tone Grid Current
Single -Tone Driving Power
Driving Impedance
Load Impedance
Plate Input Power
Plate Output Power
2000
60
265
55
100
38
160
3950
530
325
D.c. Plate Voltage
D.c. Plate Voltage
4 -250A)
volts
ma.
ma.
mo.
ma.
watts
ohms
ohms
watts
watts
4 -1000A
Zero -Signal Plate Current
Single -Tone Plate Current
Single -Tone Screen Current
Single -Tone Grid Current
Single -Tone Driving Power
Driving Impedance
Load Impedance
Plate Input Power
Plate Output Power
switch is rotated to the 80 meter position,
contact is made between the two spring arms
through the body of the screw, which completes the circuit between the switch contacts.
Amplifier
Typical operating conditions
for various tubes are tabulated
in figure 40. For initial adjustment, four or five hundred volts plate
potential is applied to the amplifier, and
sufficient grid drive is supplied (five watts
Adjustment
70
110
volts
ma.
ma.
ma.
ma.
watts
ohms
ohms
watts
watts
or so) to provide an indication on the plate
meter. The loading capacitor is set at maximum capacitance, and the tuning capacitor is
adjusted for resonance, which is indicated by
the customary dip in plate current. After resonance is found full plate voltage should be
applied to the amplifier, and resting plate
current compared with the value shown in the
table. If all is well, a carrier is applied to the
amplifier for adjustment purposes. The signal
may be generated by carrier injection, or by
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HANDBOOK
4 -400A
tone modulation of a sideband exciter.
Caution! Do not apply full excitation to
any grounded grid amplifier without plate
voltage on the stage, or with the stage improperly loaded. Under improper conditions,
driving power normally fed to the output circuit becomes available to heat the control grid
of the tube to excessive temperature, and such
action can destroy the tube in short time.
Adjustable control of the excitation level is
mandatory.
The amplifier is now loaded to full, single
tone input. (In the case of two 4- 400A's this
will be 3000 volts at 333 ma., 2500 volts at
400 ma., or 2000 volts at 500 ma.) Driving
power will be approximately 30 watts per
tube. Under these conditions, power input will
be 1000 watts p.e.p. for sideband operation.
To properly load the amplifier for 2 kw.
p.e.p, operation it is necessary to have a
special test signal. Tuning of this (or any
other linear amplifier) is greatly facilitated
by the use of an oscilloscope and envelope
detectors. Even with two -tone or carrier input
signal, however, it is difficult to establish the
proper ratio of grid drive to output loading.
In general, antenna coupling should be quite
heavy: to the point where the power output
of the amplifier has dropped about two percent. This point may be found by experiment
for power levels up to 1 kw. p.e.p. However,
since neither this amplifier, nor most power
supplies, are designed for continuous carrier
service at two kilowatts and since this average
power level is illegal, some means must be
devised to tune and adjust a "legal" two
kilowatt p.e.p. linear amplifier without exceeding the limitations of the amplifier, and
without breaking the law. A proper test signal
having high peak to average power ratio will
do the job, permitting the amplifier to run
at less than a kilowatt d.c. input while allowing the 2 kw. peak power level to be reached.
This type of signal can be developed by an
audio pulser, such as was described in QST
magazine, August, 1947 ( figure 41) The
duty cycle of this simple pulser is about 0.44.
This means that when the amplifier is tuned
up for a d.c. indicating meter reading 800
watts, using the pulser and single tone audio
injection, the peak envelope power will just
reach the 2 kw. level. An oscilloscope and
.
Amplifier
643
6J5
0010
AUDIO
INPUT
PULSED AUDIO
OUTPUT
MEA
/r.P-3045
OR
FOUI VALENT
Figure 41
AUDIO PULSER FOR HIGH POWER
TUNE -UP OF AMPLIFIER
This simple audio pulser modifies the audio
signal to the sideband exciter so that it has a
high peak -to- average power ratio. Amplifier
may be thus tuned for two kilowatt p.e.p.
input without violating the one kilowatt
maximum steady state condition.
audio oscillator are necessary for this test,
but these are required items in any well
equipped sideband station. Loading and drive
adjustments for optimum linearity consistent
with maximum power output may be con ducted by this method.
29 -10
A Pi- Network
Amplifier for C -W,
A -M, or SSB
This all- purpose amplifier covers the 3.529.7 Mc. range, and is designed for one kilowatt c.w. or s.s.b. operation, and 825 watts
input plate modulated a.m. service. Using a
single 4 -400A tetrode tube, this grid- driven
amplifier may be driven by an exciter having
a power output of approximately 15 watts.
Two mechanical designs are discussed, one
using variable vacuum tuning capacitors, and
the other employing the less expensive variable air capacitors. The latter design is highly
recommended as an inexpensive and foolproof
amplifier for the amateur wishing to go high
power on a lean purse!
Amplifier Circuit
The schematic of the amplifier is shown in figure
43. Bandswitching is employed in the grid
and plate circuits, and the tetrode tube is
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644
H.F.
THE RAD
Power Amplifiers
Figure 42
4 -400A ALL -BAND AMPLIFIER
This compact amplifier is designed for operation in the 3.5 -29.7 Mc. ronge. Using bandswitching in
the grid and plate circuits, the unit is capable of a full kilowatt input on c.w. and s.s.b., and 825
watts a.m. phone. The amplifier employs variable vacuum tuning capacitors, but an alternative
design uses inexpensive air capacitors. Panel controls are (I to r.): plate current meter (top), grid
bandswitch (center), and grid tuning (bottom). Screen current meter (top), plate tuning (center),
and plate loading (bottom). Grid current meter (top), plate bandswitch (center), and filament
switch and pilot lamp (bottom).
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;O
HANDBOOK
4 -400A
Amplifier
645
EA
300
,0 KV
4- 250A /4 -400A
RF
L 2
OUT.
RFC
ISO
CS
2
470
S1A
Ext.
1500
S4
L1
27.1V.
60
SOO
SEE
RFC,
NOTE
110MV
NOTE.
C IS .01
Ur
CORAM IC, 600 VOLT.
Hn
B+HV
SCLr MODULATION CIRCUIT
OC
FOR SCRCCN LOAD.
X
OMONE
12 ,011
zwll
--IHII
470
C
-BIAS
B
MV
115V."1.
Uri
O
+SCR
CND.
+BIAS
-SCR.
Figure 43
SCHEMATIC, 4 -400A AMPLIFIER
C1 -140 ppfd. Hammarlund
C2- Neutralizing capacitor.
APC -1408
10 µpfd. Millen
#15011, or Johnson N -250
UCS
-250 variable
Jennings
C3 -250 µµid.
vacuum capacitor. Johnson 250070(153 -13)
C4 -1500 ppfd. Jennings UCSL -1200 variable
vacuum capacitor. J. W. Miller #2I04 air
capacitor may be substituted
L1 -50 turns, #24, 13/4" long, 3/4" diam. Tap
S, 8, 13, and 25 turns from grid end.
Wound on ceramic form. Link coil is 4 turns
#18 insulated wire, wound on "cold" end
of LI, tapped at center of winding
L2- Barker d Williamson #850 pi- network inductor. 80 meters, 13.5 ph.; 40 meters, 6.S
ph.; 20 meters, 1.75 ph.; 15 meters, 1.0 ph.;
10
meters, 0.8 ph.
-0 -50 d.c. milHammeter
M2 -0 -100 d.c. milliammeter
M3 -0 -800 d.c. milliammeter
PC-4 turns, 1" diam. wound about
MI
four 220
ohm, 2 watt composition resistors in parallel
RFC,,s -2.5 mh. National R -100
RFC2 -BSW #800 plate choke, or National
R -175A
S1
pole, S position ceramic switch. Centralab 2002
volts @ 15 amperes. Triad F -9U
T1
Blower-Shaded pole induction motor, 2400
r.p.m. 4 blade fan, 21/2" diam. Allied Rodio
Co., Chicago, part number 72P -715
Counter dials: Grath Mfg. Co.
-2
-5
neutralized to achieve maximum stability of
operation. Link coupling from the external
exciter is used, and a tuned grid circuit offers
maximum rejection to any spurious harmonics
or unwanted emissions of the exciter. Capacitive bridge neutralization is employed, with
a 250 ¡yid. mica capacitor forming the
ground leg of the bridge in the grid circuit.
Each screen terminal of the tube socket is
bypassed to ground with a low inductance
high voltage ceramic capacitor, and the screen
power lead is harmonic filtered by a simple
R -C network. Grid and screen currents are
separately metered. To aid circuit stability in
the region of v.h.f. parasites, one leg of the
filament is grounded, and the opposite terminal is bypassed to ground at the tube socket.
In addition, simple parasitic chokes are used
in the grid and plate circuits as a safety
measure. The plate circuit is the popular pinetwork configuration, and will match 50or 75 -ohm antenna loads having an s.w.r. of
less than 2 to 1.
Amplifier plate current is metered in the
B -minus lead to the power supply in order
to remove the meter from the high potential
B -plus circuit. By returning the bias and
screen supplies to the cathode circuit (ground)
the plate meter reads only the true plate current and not the cathode current, which is
the sum of grid, screen, and plate currents.
The reader is referred to the discussion of
this subject in a previous section of this
chapter.
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THE RADIO
H.F. Power Amplifiers
646
Figure 44
TOP VIEW OF 4 -400A AMPLIFIER
R.f. circuits atop the chassis are enclosed in ventilated box made of perforated aluminum. Band switching inductor is at the right, with coaxial antenna receptacle directly to the rear, mounted
on aluminum plate. To left of variable vacuum capacitor is the disc -type neutralizing capacitor.
Plate r.f. choke is directly behind tube. Panel meters are isolated from r.f. field by aluminum
sub -panel.
Amplifier
The amplifier is constructed
upon an aluminum chassis
measuring 15" x 17" x 4 1/2".
Standard, TVI -proof construction is used, as
outlined in the Workshop Practice chapter
of this Handbook. The above -chassis circuitry is enclosed in a perforated aluminum
Construction
enclosure measuring 13 t/4" x 17" x 9 ". The
frame of the enclosure is made of t/2 -inch
aluminum angle stock, with corner gusset
plates. Perforated sheets form the sides and
top and are held in position with sheet metal
screws spaced about three inches apart along
the edges of the material. A sub -panel made
of I/8-inch aluminum is placed about 13/4
www.americanradiohistory.com
HANDBOOK
4 -400A
inches behind the main panel. The area between the two panels is taken up by the three
meters, and the gear drive system for the grid
bandswitch. The panels are held in position
by metal spacers located at the extreme top
corners of the assembly.
Placement of the major components may
be seen in the photographs. The pi- network
tuning capacitors are centered on the panel,
with the bandswitch controls placed symmetrically about the tuning capacitor. Below deck
the output loading capacitor is contained
within a small shielded compartment formed
from sheet aluminum. As the grid input circuit is adjacent to this capacitor, it is important that :here be no leakage of r.f. energy
from input to output circuits. The bottom
plate of the chassis is a solid piece of aluminum, with a 4 -inch hole cut in it directly
below the blower for the tube socket. The hole
is covered with perforated aluminum stock,
and the bottom plate is firmly bolted to the
chassis lip, and also to the flanges of the box
screening the output loading capacitor. An
"r.f.- tight" box thus surrounds the capacitor.
Connection between the capacitor and the pinetwork circuit above the deck is made via a
ceramic feedthrough insulator mounted in
the deck.
The blower motor is mounted in a vertical
position below the ceramic tube socket (figure
44A ). A strip of aluminum supports the motor
between the lip of the chassis and a lip of
the capacitor compartment. The bracket is
mounted with flat-head bolts, and the motor
bolts are run through rubber grommets
mounted in the strip. The power leads to the
motor, as well as all other low voltage power
wiring beneath the chassis, are run in shielded
braid with the lead bypassed to the braid at
each end of the run.
The grid circuit components are mounted
to an aluminum plate spaced away from the
panel by four aluminum posts. The grid capacitor is driven by two flexible couplings from
the tuning dial, which is positioned on the
panel below the bandswitch and meter. The
grid bandswitch is driven from atop the
chassis by means of two right -angle gear
drives. One drive is below the chassis and
the second is placed in the meter compartment
behind the bandswitch dial.
Amplifier
647
Placement of the major plate circuit components may be seen in figure 44. The
tuning capacitor is centered on the chassis with
the tube and neutralizing capacitor on one
side, and the plate tank inductor on the opposite side. The ceramic plate circuit coupling
capacitors are mounted between two aluminum
plates, forming a "sandwich" supported on one
side by a 1/2-inch wide copper strap from the
plate r.f. choke, and on the other side by a
similar strap affixed to the plate tank
capacitor.
The bias and screen supply
described in the next section of this chapter may
be used for all- purpose amplifier operation.
Screen protective relay RY1 should be adjusted
to cut out at a maximum screen current of
50 milliamperes. If sideband operation is not
contemplated, it is possible to eliminate the
voltage regulator tubes in the screen supply
and substitute a simpler unit that will provide 400 volts d.c. at 50 milliamperes. This
will be suitable for either phone or c.w. operation. For the former, it is necessary to allow
the screen to "self- modulate" itself to obtain
100 percent plate modulation. This is done
by inserting a 10 -henry filter 100 ma. choke
in the screen lead at the point marked "X"
(figure 43) . The choke is shorted out for
c.w. operation.
Bias and Screen
Supply
Use of
Air
In order to reduce the cost of
the amplifier, it is possible to
substitute air capacitors for the
variable vacuum units. A Johnson #250D70
(153 -13) will serve as the plate capacitor,
and a four gang b.c. -type capacitor, such as
the J. W. Miller #2104 will replace the
vacuum output capacitor. In addition, the inexpensive Air -Dux inductor and the ceramic
switch described in the "KW -2" amplifier
may be used as a substitute for the more
expensive bandswitch assembly shown here.
Capacitors
Amplifier Tuning
and Adjustment
The amplifier should be
neutralized in the manner
described in the next section of this chapter. Proper neutralization is
indicated during operation of the amplifier
by detuning the plate tuning capacitor a small
amount each side of resonance. The point of
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648
H.F. Power Ampli=fiers
THE RADIO
Figure 44A
LAYOUT OF UNDER -CHASSIS COMPONENTS
The pi- network loading capacitor is mounted on angle plates within the shielded compartment at
center. The grid circuit components a-e at the left, in fient of blower fa- and motor. The filament
transformer is mounted to the wall at right side of chassis. Shielded wire .s used for all low- voltage
power leads.
www.americanradiohistory.com
HANDBOOK
Kilowatt Amplifier
minimum plate current should coincide with
the point of maximum grid current. If grid
current increases when the plate circuit is
tuned either side of resonance, the setting of
the neutralizing capacitor should be varied
slightly until the two readings coincide at one
capacitor setting.
The bias supply is adjusted to provide approximately -120 volts of cut -off bias. Full
screen voltage may be applied as long as cutoff bias is on the stage. Full excitation, however, should never be applied in the presence
of screen voltage unless full plate voltage is
on, and the amplifier is properly loaded.
Screen current is a very sensitive indicator of
proper operation. High values of screen current point to insufficient antenna loading, or
to excess drive. Low screen current indicates
excessive antenna loading or insufficient drive.
If the plate current seems normal, the drive
level should be adjusted to provide proper
screen current.
29 -11 Kilowatt Amplifier
for Linear or Class C
A pair of 4 -250A or 4 -400A tetrode tubes
may be employed in a pi-coupled amplifier
capable of running one kilowatt input, c -w or
plate modulated phone, or two kilowatts
p.e.p. for sideband operation. Correct choice of
bias, screen, and exciting voltages will permit
the amplifier to function in either class A, B,
or class C mode. The amplifier is designed
to operate at plate potentials up to 4000 volts,
and excitation requirements for class C operation are less than 25 watts.
A bandswitching type of pi- network is employed in the plate circuit of such an amplifier, shown in figure 45 The pi- network is
an effective means of obtaining an impedance
match between a source of r.f. energy and a
low value of load impedance. A properly designed pi- network is capable of transformation
ratios greater than 10 to 1, and will provide
approximately 30 decibels or more attenuation
to the second harmonic output of the amplifier
as compared to the desired signal outpiat. Since
the second harmonic level of the amplifier
tube may already be down some 20 db, the
actual second harmonic output of the network
will be down perhaps 50 db from the fundamental power level of the transmitter. Attenuation of the third and higher order harmonics
will be even greater.
Operation
Figure 45
GENERAL PURPOSE
AMPLIFIER
OPERATES IN CLASS
A, 8, OR C MODE
This
kilowatt
employs
a
amplifier
pair
649
of
4- 250A's in a pi- network
circuit. Mode of opera-
tion may be set by selection of proper screen and
bias voltages. Grid, plate,
and screen current meters are
mounted on
plastic plate behind panel
cut -out, and tubes are
visible through shielded
panel opening. Across
bottom of panel (left to
right' ore bandswitch,
grid tuning, plate tuning, loading, and primary
power control
circuits.
Plate tuning knob is attached to small counter
dial.
www.americanradiohistory.com
650
H.F. Power Amplifiers
Le
4 -250A
4 -250A
4 -4004
4 -400A
L2 L3
L
EACH
001
5NV
S
S
OUTPUT
1
J2
H(
L4 Ls
RFCI
RFC 2
5ó1 1
500
20NV
NOTE: SCREEN BYPASS CAPACITORS
ARE CENTRALAB TYPE 838
INPUT
JI
01
/M3
y
01
+T
Hl1
L
TS1
-BIAS
CON-
rROL
115V
+SCREEN GN0
ti
115v.
B+2500-
ti
3500
Figure 46
SCHEMATIC, GENERAL PURPOSE KILOWATT AMPLIFIER
S -Two pole, 6 position
ohm, 2 watt
PC-47
T -S volt, 20 ampere.
-100
µp /d. Hammarlund HF -100
Cr-200 µµId., 10KV variable vacuum capacitor. Jennings UCS -200
Cs-1500 µtd., variable
Cardwell
capacitor.
CI
8013
C.-Neutralizing capacitor, disc. Millen 15011
C1
-300
µµtd.,
1250 volt
L1 -L,,
-See
coil
mica,
table
composition
wound with
6
switch. Two Centralab
PA -17 decks, with PA301 index assembly
resistor
turns
= 18e.
RFC:
-2.5
mh.
choke.
R -100
National
-Two
S:
RFC.-Heavy
duty, wide band r.i. choke. Barker 8 Williamson type
800
RFC
-VHF
ite
51
choke. Ohm-
Z -144
The peak voltages encountered across the
input capacitor of the pi- network are the same
as would be encountered across the plate tuning capacitor of a single -ended tank used in
the same circuit configuration. The peak voltage to be expected across the output capacitor
of the network will be less than the voltage
across the input capacitor by the square
root of the ratio of impedance transformation
of the network. Thus if the network is transforming from 5000 ohms to 50 ohms, the ratio of impedance transformation is 100 and
the square root of the ratio is 10, so that the
voltage across the output .capacitor is 1 /10
that across the input capacitor.
A considerably greater value of maximum
capacitance is required of the output capacitor
than of the input capacitor of a pi- network
when transformation to a low impedance load
is desired. For 3.5 Mc. operation, maximum
values of output capacitance may run from
Stancor
M:
pole, 6 position
high voltage switch.
Communication Products Co. type 88 two
gang switch
-Four
pole, three position switch. Centra lab
P -6492
-0
50
Triplett
-
-
150
Triplett
ma.
ma.
d.c.
d.c.
-0
750
ma.
d.c.
Triplett
Gears-2 required. Boston Gear CG -465 and
M
-
=G-466
500 µµfd. to 1500 µµfd., depending upon the
ratio of transformation. Design information
covering pi- network circuits is given in an earlier chapter of this Handbook.
Illustrated in this section is an up -todate version of an all -band pi- network amplifier, suited for sideband or class -C operation.
The unit is designed for TVI -free operation
over this range.
Circuit
The schematic of the general
purpose amplifier is shown in
figure 46. The symmetrical
panel arrangement of the amplifier is shown
in the front view (figure 45) and the rear
view (figure 47) . A 200 µµfd. variable vacuum capacitor is employed in the input side
of the pi- network, and a 1500 µµfd. variable
air capacitor is used in the low impedance output side. The coils of the network are switched
in and out of the circuit by a two pole, five
Description
www.americanradiohistory.com
Figure 47
REAR VIEW OF GENERAL PURPOSE
AMPLIFIER WITH SHIELD
REMOVED
The pi- network circuit is built from an inex-
pensive high voltage rotary switch, and five
inductors. The switch is panel driven by a
gear and shaft system shown in figure 38.
Variable vacuum capacitor is mounted vertically between the tubes, directly in back of
the plate r.f. choke. Neutralizing capacitor is
at right, connected to plates of tubes with a
wide, silver plated copper strap. Meters are
enclosed by aluminum shield partition running
the width of the enclosure, with conduit carrying meter loads to under- chassis area at
left, front of chassis. Metal shells of tube
bases are grounded by spring contacts.
position high voltage ceramic rotary switch.
Each coil is adjusted for optimum circuit Q,
resultine in no tank circuit compromise in efficiency at the higher frequencies. A close -up
of the tank circuit is shown in figure 47. The
plate blocking capacitor is made of two .001
µfd., 5 kv. ceramic capacitors connected in series.
Special precautions are taken to insure operating stability over the complete range of
amplifier operation. The screen terminals of
each tube socket are jumpered together with
Ye" copper strap and a parasitic choke (PC)
is inserted between the center of the strap and
the screen bypass capacitor. In addition, sup-
O
u
pressor resistors are placed in the screen leads
after the bypass capacitor to isolate the sensitive screen circuit from the external power
leads. A third parasitic choke is placed between
the grid terminals of the tubes and the tuned
grid circuit.
The five coils of the grid circuit are enclosed in a small aluminum shield placed adjacent to the tube sockets (figure 48 and figure
50). The amplifier is neutralized by a capacitive bridge system consisting of neutralizing
Figure 48
PLACEMENT OF
PARTS IN UNDER CHASSIS AREA
Grid tuned circuit is en-
closed in separate enclosure at left. Bandswitch
projects out the rear of
case, and is gear driven
by same shaft that actuates the plate band switch. Switches are driven through right -angle
gear drives and gears.
Output capacitor of pinetwork is shielded from
under -chassis
of
rest
components.
The screen terminals of
each tube socket are
strapped together with
ribbon, and
3/e" copper
inductance screen
low
is
capacitor
bypass
socket
to
grounded
mounting bolt. Screen
parasitic choke mounts
between strap and ca-
pacitor
terminal.
All
power leads beneath the
chassis are run in shield-
ed braid, grounded to
at convenient
chassis
points. B -plus lead is
made of section of RG8,'U coaxial cable, with
outer sheath and braid
removed.
www.americanradiohistory.com
H.F. Power Amplifiers
652
COIL TABLE
T H
FIGURE 49
KILOWATT AMPLIFIER
FOR
GRID COILS
Li-
(SO METERS)
:
L2 -(40 METERS I:
L3-(20 METERS):
N24 E, 3/4' O /A., 1" LONG ON
AMPNENOL POLYSTYRENE FORM.
40 TURNS
24
3O TURNS
E, .7/4"DIA., 3/4- LONG
ON AMPNENOL POLYSTYRENE FORM.
72 TURNS,
82W 3071 MINIDUCTOR,
L4 -(1S METERS): 7TURNS,
88 3O1O MINIDUCTOR,
S/4" OM., 3/4'
LONG.
3/4' DIA., 7/e' LONG.
LE
-(10 METERS):
3
TURNS. AS ABOVE.
ALL COILS HAVE 3 TURN LINKS MADE
OF HOOKUP
WIRE.
PLATE COILS
LS-
METERS)
77 TURNS
!O, 3
L7- (40 METERS):
IO TURNS
IO,
L! -lao METERS):
P
(SO
:
-O
O., 4
TURNS PER INCH
-AIR -DUX
LP
-(1S METERS):
L10 -00 METERS):
TURNS,
2
1/2'
3/7"
O.O.,
3" OD.,
COPPER TURING,
LONG.
1/4"
COPPER TURING.
/4-
COPPER TURING.
7
TURNS,
1/4" 0.0., .1" LONG.
S
TURNS, I
I/4" 0.0.,
-
TURNS PER /NCH
3"
2
2
5
3
"LONG.
capacitor G and Cr., the grid circuit bypass capacitor.
Screen voltage may be removed for tune -up
purposes by control switch 5., section B. The
screen circuit is grounded in the "off" and
"fil" positions by means of switch section C.
Amplifier
The complete amplifier is
built upon an aluminum chassis measuring 13" x 17" x 3"
and has a 14" standard relay rack panel. The
Construction
Figure 50
GRID TANK CIRCUIT ASSEMBLY
Coils are mounted to the ceramic switch decks
by their leads. A small aluminum plate attached to rear of the switch assembly rods
supports grid tuning capacitor which projects
out rear of shielded enclosure. Entire assembly
may be pre -wired before placing in enclosure.
E
R
A
D
I
O
grid circuit components are mounted within
an aluminum box measuring 3" x 4" x 4 ".
Plate loading capacitor G, r.f. choke RFC -1,
and output connector J2 are placed within an
enclosure measuring 6" x 6" x 3", made up
of aluminum angle sections and sheet material.
The plate circuit shielding is made of Reynold's
"Do- it- yourself" aluminum stock, available at
most hardware stores.
Layout of the major components can be seen
in figure 47. The two tube sockets are placed
directly behind the panel opening, with the
plate r.f. choke between them, and the variable
vacuum capacitor is mounted vertically to the
chassis directly behind the sockets, on the center line of the chassis. To the right of the sockets is neutralizing capacitor C.. The high voltage ceramic coil switch SA-B is placed directly
behind the vacuum capacitor, mounted in a
vertical position.
The variable vacuum capacitor is panel driven by a counter -type dial, through a miniature
right angle gear drive, as seen in the under chassis view (figure 48). The plate and grid
band switches are ganged and switched in unison by means of a shaft acting through two
right angle gear drives and two bevel gears.
Both circuits are thus switched by the "Band switch" control located in the lower left corner
of the front panel.
It is necessary to apply forced air to the
sockets of the amplifier tubes. A large 115
volt a.c. operated blower is therefore mounted
in the center of the bottom shield plate. The
under -chassis area is thus pressurized and the
majority of the air escapes through the socket
ventilation holes located near the pins of the
tubes.
All wiring beneath the chassis (with the
exception of the filament leads) is done with
5KV insulated wire, encased in metallic braid
which is grounded to the chassis every inch or
so. The B -plus wiring from the high voltage
terminal to the plate current meter is done
with a section of RG -8 /U coaxial line from
which the outer braid has been removed. A
similar piece of line is run from the ,peter to
the plate r.f. choke, RFC -2.
The three meters are mounted upon a lucite sheet placed behind a second lucite sheet
mounted behind a cut-out in the front panel.
The meters are shielded from the plate circuit
of the amplifier by an aluminum enclosure
that covers the wiring and meters, running the
full length of the chassis. The meter leads pass
through the plate circuit area via a short length
of 1,1,-inch aluminum conduit that is threaded
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HANDBOOK
Kilowatt Amplifier
4-250A /4-400A OPERATING CHARACTERISTICS
(2 TUBES
)
MODE
ITEM
PLATE VOLTAGE
55861
33682
PHONE
C.W.
3000
3000
2500
3000
400
330
600
S00
400
500
24
1.0
60
70
GRID BIAS
-110
-60
PROTECTIVE BIAS
-110
-60
-200
-120
-200
-120
PLATE CURRENT (MA.)
110
SCREEN VOLTAGE
SCREEN CURRENT (MA.)
GRID CURRENT (MA.)
POWER OUTPUT(WATTS)
Ti
-420 260 -440
0
0
20
20
600
700
770
600
5R4 -GY CH,
R1
5K
sow
RY1
111
2
o
1
loF
nv
5
T
U
VR -1150
5GR
5
VR -150
1.3 v.
1oó
VR
1s
OR
V
115V.
R- 150
VR
T2
/W
ID
-90
9
GND
5
OR
VR -150
106.3V.
`\\\
V.
(
((
5Y3-GT
*
20LIF
I
450
5K/25w
R3
+
R4
R2
6
I
LW
e
St
-BIAS
Figure 51
OPERATING DATA AND SCHEMATIC,
SCREEN AND BIAS SUPPLY
volts of 150 ma. and
-0- 410 -870
T:- 870 -410
volts,
ma.
60
5
Stancor
P
Te- 235 -0 -235
CH
CH
2
o.,
6.3
v.
3.5
a.
-8307
volts at 40 ma. Stanco, PC-
8401
-7
henry
-7 henry
at 150 ma. Stancor C -1710
at 50 ma. Stancor C -1707
relay, adjustable 100 -250 ma.
insulated from chassis.
RY,- Overload
Note:
J, is
at each end and bolted to the chassis and the
meter shield. Plate circuit wiring above the
chassis is done with 1/2-inch silver plated copper strap.
After the amplifier is wired
and checked, it should be
neutralized. This operation
can be accomplished with no power leads attached to the unit. The tubes are placed in
their sockets, and about 10 watts of 30 Mc.
r.f. energy is fed into the plate circuit of the
amplifier, via the coaxial output plug J,. The
plate and grid circuits are resonated to the
Amplifier
Neutralization
653
frequency of the exciting voltage with the aid
of a grid -dip meter. Next, a sensitive r.f. voltmeter, such as a 0 - 1 d -c milliammeter in
series with a 1N34 crystal diode is connected
to the grid input receptacle (J1) of the amplifier. The reading of this meter will indicate
the degree of unbalance of the neutralizing
circuit. Start with a minimum of applied r.f.
excitation to avoid damaging the meter or the
diode. Resonate the plate and grid circuits for
maximum meter reading, then vary the setting
of neutralizing capacitor G until the reading
of the meter is a minimum. Each change in G
should be accompanied by re- resonating the
grid and plate tank circuits. When a point of
minimum indication is found, the capacitor
should be locked by means of the auxiliary set
screw.
Complete neutralization is a function of the
efficiency of the screen bypass system, and
substitution of other capacitors for those noted
in the parts list is not recommended. Mica,
disc -type, or other form of bypass capacitor
should not be substituted for the units specified, as the latter units have the lowest value of
internal inductance of the many types tested
in this circuit.
The amplifier requires -60 to
-110 volts of grid bias, and
plus 300 to 600 volts of
screen potential for optimum characteristics
when working as a class AB1 linear amplifier. Screen voltage for class C operation
(phone) is 400 volts. The voltage may be
raised to 500 volts for c.w. operation, if desired, although the higher screen voltage does
little to enhance operation. Approximately -120
volts cut -off bias is required for either phone
or c -w operation. A suitable bias and screen
power supply for all modes of operation is
shown in figure 51, together with an operating
chart for all operating voltages. The supply
furnishes slightly higher than normal screen
voltage which is dropped to the correct value
by an adjustable series resistor, R1. This series
resistor is adjusted for 30 milliamperes of current as measured in meter jack J1 when the
supply is disconnected from the amplifier.
Series bias resistor R2 is adjusted for the same
current in jack J. under the same conditions.
The value of protective bias may now be set
by adjusting potentiometer R3.
Additional bias is required for class C operation which is developed across series resistor
R.. Switch S1 is open for class C operation and
closed for sideband operation.
It is imperative that the screens of the tetBias and
Screen Supply
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654
H.F. Power Amplifiers
rode amplifier tubes be protected from excessive current that could occur during tuning
adjustments, or during improper operation of
the amplifier. The safest way to accomplish
this is to include an overload relay that will
open the screen circuit whenever the maximum
screen dissipation point is reached. Two 4250A tubes or 4 -400A tubes have a total screen
dissipation rating of 70 watts, therefore relay
RY-1 should be adjusted to open the screen
circuit whenever the screen current reaches
approximately 100 milliamperes.
29 -12
A 2- Kilowatt
All -Band
P.E.P.
Amplifier
T H
E
R
A D
I
watts p.e.p. output in sideband service. Maximum grid dissipation of the 4CX1000A is
zero watts. The design features which make
the tube capable of maximum power operation without driving the grid into the positive
region also makes it necessary to avoid positive
grid operation.
This efficient amplifier covers the 3.5 -29.7
Mc. amateur range and may be driven by any
modern sideband exciter having a power output of 75 watts, p.e.p. In addition to sideband
operation, the amplifier may be used as an
a.m. linear, providing a carrier power of about
350 watts.
Circuit Description
Described in this section is a deluxe all band linear amplifier suited for s.s.b. or c.w.
operation up to the maximum legal power
limit. A 4CX1000A ceramic power tetrode is
employed in a basic passive grid circuit shown
earlier in this chapter in figure 11C.
The 4CX1000A is a ceramic and metal,
forced air -cooled, radial beam tetrode with a
rated maximum plate dissipation of 1000
watts. It is a medium voltage, high current
tube specifically designed for Class AB1 r.f.
linear amplifier service where its high gain
and low distortion characteristics may be used
to advantage. At the maximum rated plate
voltage of 3000, the tube is capable of 1680
O
The circuit of this all band amplifier is shown
in figure 53. A resistance loaded, passive grid
configuration is employed in conjunction with
a pi- network output circuit. Grid drive requirements are about 60 volts peak, developed
across resistor R1 which has a value of 50
ohms. This corresponds to approximately 72
watts p.e.p., all of which is dissipated in the
grid resistor. Average power dissipated in this
resistor is about 30 watts under voice waveform conditions. It is possible to tune up and
adjust the exciter with the plate and screen
voltages removed from the 4CX1000A, using
this resistor as a dummy load.
The amplifier plate circuit is the popular
pi- network configuration employing a tapped
Figure 52
DELUXE 4CX1000A
SIDEBAND
AMPLIFIER
Constructed in a desktop cabinet, this 5 -bond
sideband
amplifier is
rated at 2 kilowatt p.e.p.
level. Panel controls are
(I. to r.): meter switch,
plate tuning (top), filament and plate switches
and pilots (center), plate
loading control (bottom)
and bandswitch.
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HANDBOOK
4CX 1000A Amplifier
EA
4CX1000A
655
500
RF OUT
RF
.001
zKV
IN
1
R FC 2
=
0.1
tKx42
500.
SEE TEXT
W
E'
E
R
ADJ.
B
5000
/AS
2w
15+ HV
R F
OUTPUT
E'
rDW
,0K
MULTIMETER SWITCH S1
S,*
,ow
A
- 0 -, MA., GRID
BSIB
C
l
0 -1
- 0
MA., PF OUT.
-40
MA
,
SCREEN
D- 0- 500V.,
111,
SCREEN
e-0 -5KV.,
DA
PLATE
0 -1
NOTE
2
-BIAS
A.C.
PIL
B+
HEUT
SW
ON
5V.
N
B
B+
-500
µµtd.,
5
kv. Jennings Radio Co., type
UCSL
-2000
NF,td., 2 kv. Jennings Radio Co.,
type UCSL
8 Williamson x.852 turret
RFC,-2.5 mh. National R -100
RFC2- Transmitting type r.f. choke. National
R -175A
T, -6.0 volts @ 11 a. Stancor P -6463
Blower -S0 cu. ft. min. Ripley »81 or equivalent
C2
80P
GND.
SCR.
Figure 53
SCHEMATIC, 4CX1000A AMPLIFIER
C1
B-
0.1 CAPACITORS ARE SPRAGUE
-3
NYPASS
L,- Barker
coil and variable vacuum capacitors. A simple
diode voltmeter is used to monitor the r.f.
output voltage of the amplifier. The network
is capable of matching antenna loads of 50 -75
ohms, which exhibits an s.w.r. of less than 2/1.
Metering and Control Circuits. The amplifier
unit contains two panel meters (figure 54) .
A 0 -1 d.c. ammeter placed in the B -minus leg
of the power supply serves as a plate meter.
The second meter is a 0 -1 d.c, milliameter
connected as a multi -purpose indicator. Panel
switch S1 places the meter across shunt and
multiplier resistors in various circuits.
The basic control circuits are shown in
figure 55. A "fail- safe" design utilizes control
relays energized from low voltage d.c. supplies.
If one of the supplies fails, or a relay becomes
inoperative for any reason, the 4CX1000A
tube is protected from abuse. Inexpensive 115
volt a.c. relays are used, which operate satisfactorily from a d.c. source of about 30 volts.
Series resistors may be used with the relay
coils to provide the correct pull -in voltage.
Relay RY, is the main power relay. When the
"Filament On" switch on the amplifier is
thrown, the bias supply ( -150 volts) is energized. Power is applied to relay RY1 through
the overload relay contacts (RY_B) and the
time delay relay, TD. For usual voice operation, the plate supply is left on at all times.
Relay RY1 may be released by the overload
relay RY., whose actuator coil is placed in
series with the B -minus lead of the high voltage supply. The overload relay is adjusted to
trip at a plate current of approximately 850
ma. Once the relay is tripped, it is reset by
the auxiliary reset coil by momentarily throwing off the filament switch.
Screen voltage of the 4CX1000A is obtained from the high voltage supply through
www.americanradiohistory.com
656
H.F. Power Amplifiers
an adjustable dropping resistor and is controlled by two OD3 regulator tubes. With this
regulator and divider combination, the screen
voltage is stabilized at 300 volts, yet a maximum of only 10 watts may be drawn from the
supply. This protects the 4CX 1000A under any
operating conditions, as the maximum screen
dissipation rating is 12 watts. In the event the
plate supply fails, the tube is protected from
screen overload, as the screen voltage is also
removed at the same time. In case of bias
failure, the plate circuit relay is de- energized,
as it receives power from the bias supply.
The screen current of the 4CX1000A varies
over a wide range, depending upon the tube
operating conditions, and may approach
Figure 54
HINGED FRONT PANEL REVEALS
BIAS AND VOLTMETER
ADJUSTMENTS
The main panel is hinged at the lower edge,
and is cut out to permit meter switch and
band switch knobs to project through. Special
dial plates are cut from lucite and engraved
for the two controls. At left, the two control
potentiometers are mounted below meter
switch.
T H
E
R
A D
I
O
moderate negative values if the tube is lightly
loaded. It is convenient, therefore, to be able
to monitor negative values of screen current.
A bleeder resistor is placed directly at the tube
socket after the screen meter shunt. With 300
volts applied to the screen, this resistor is
adjusted to provide a static reading of 20
milliamperes on the meter. Thus, 20 ma. must
be subtracted from the meter reading to obtain the actual screen current. When the screen
current is negative, the meter reading will drop
below 20 milliamperes. A negative screen
current of 18 ma., for example, will result in
an indication of plus 2 ma. on the meter.
Negative screen currents as great as -20 ma.
can be monitored in this manner.
The amplifier is cut off during standby
periods by means of relay RY3, which boosts
the grid bias to -150 volts. This prevents the
amplifier from generating troublesome diode
noise during periods of reception. Full operating potentials are applied to the amplifier at
all times, and the amplifier is activated by removing the blocking bias. Relay RY3 may be
controlled by an external voice circuit; it is
only necessary to ground pin #2 to pin #3 on
the control strip (figure 55) to activate the
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HANDBOOK
C.C.
Ow--471---1
ORY,eI
I
0
'
42 01
-
O
250
3000 V.
I
100
SUPPLY
I
115 V
I
I
W
300
SEE
TEXT
RYI
ADJUST
V.
V.
OD3
OVERLOAD
RY2e
0+3000
251c
EA 200w
42301
A
657
RYIA
20A.
115V
4CX1000A Amplifier
RY2
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RY2
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TO CONTROL SW.,
ANT. RELAY, ETC.
Figure 55
SCHEMATIC, AMPLIFIER CONTROL CIRCUIT
RY1- Primary control relay. 115 volt a.c. coil or d.c. coil chosen to work with bias supply voltage.
13 ampere contacts. Potter it Brumfield type PR7AY
RY3- Overload relay, l IS volt reset coil. Potter 8 Brumfield type GCIIA
RY3 -SPST, 115 volt a.c. relay. Potter B Brumfield type KLSA
SR1,2 -500 ma. rectifier. Sarkes- Tarzian M -500
T1 -150- 160 -170 volts @ 0.S amp. Triad R -93A Set at 170 volt tap
TO- Thermal
delay unit. Amperite 115 -NO -180
amplifier. The coil of the antenna relay may
be placed in parallel with that of relay RY3
for completely automatic voice operation.
Amplifier
This amplifier is an excellent
example of high -grade amateur
construction. The unit is housed
within an aluminum cabinet measuring 10"
high, 15" wide and 151/2" deep. A false front
panel, hinged at the lower edge (figure 54)
is employed for decorative purposes. An auxiliary panel is placed behind this, holding the
panel meters, control switches and the counter
dials (figure 56). This auxiliary panel is
spaced in front of the amplifier enclosure
(figure 57) . Electrical connections between
the amplifier and the auxiliary panel equipConstruction
ment are made by means of two disconnect
plugs, permitting the auxiliary panel to be
wired and tested as a complete sub -assembly,
or to be removed for servicing.
Placement of the major components within
the amplifier box may be seen in the top view,
figure 58. No chassis is used, other than two
shield boxes which enclose the tube socket and
the power receptacles on the rear of the
cabinet.
The 4CX1000A tube is mounted on the top
of an aluminum box measuring 6" square and
4" high. Only four connections pass into this
compartment: Filament, screen, and bias leads
(via coaxial capacitors seen in figure 57);
and the excitation lead (via a coaxial plug and
receptacle placed beneath the blower motor).
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658
H.F. Power Amplifiers
Power and control leads from the shielded
receptacle at the rear of the cabinet pass
through a 3/4" aluminum conduit tube to the
various circuitry mounted on the auxiliary
front panel. The high voltage lead leaves the
shield box via a short length of copper tubing to the bottom of the plate r.f. choke,
which is supported from the rear wall of the
cabinet.
The variable vacuum capacitors are mounted
to the tube socket assembly box by means of
a heavy aluminum bracket, and are driven by
the counter dials through flexible shaft
couplers.
The space between the front panel and the
enclosure is about 31/4" and the filament
transformer is mounted in the lower right portion of this area (figure 57). The enclosure
is sealed by a hinged lid, which is TVI- proofed
by phosphor- bronze finger stock fastened
around the edge of the cabinet opening.
The passive grid resistor (R1) is made up
of nine 470 -ohm, 2 -watt composition resistors
Figure 56
REAR VIEW OF METER PANEL
Counter dial mechanisms, pilot lamps, meters
and switches are mounted on aluminum sub panel. Meter switch, potentiometers, and meter
resistors are mounted on phenolic panel at
lower right. Panels have disconnect plugs so
that they may be wired separately.
T H
E
R
A D
I
O
placed in parallel (figure 59) . The resistor
leads are clipped short, and the units are
mounted between two copper discs, 11/4" in
diameter. The assembly is enclosed in a copper
tube measuring 11/2" inside diameter, 21/2"
long, and having a 1/16" wall. After the resistor assembly is completed, it is bolted to
a plate which fastens to one end of the tube.
The container is then filled with transformer
oil through a vent hole in the top. When it
is full, it is placed in a pan of water which
is heated to the boiling point. The oil expands
and drives the air out through the second vent
hole. Before the unit cools, the vent holes are
closed with solder. This compact assembly will
handle up to nearly 100 watts on an intermittent basis.
If it is desired to make a less complex resistor assembly, thirty 1500 -ohm, 2 -watt composition resistors may be connected in parallel
between two copper plates, in the manner
shown in the photograph. This arrangement is
cooled by the flow of air past the resistors.
Amplifier Adjustment Before power is applied
and Tuning
to the amplifier, filament voltage should be
adjusted to provide 6.0 volts at the socket of
the 4CX 1000A. Voltage should be held within
plus -or -minus 5 percent for maximum tube
life, so an accurate voltmeter is required for
this check.
www.americanradiohistory.com
HANDBOOK
4CX 1000A
Amplifier
659
Figure 57
MAIN PANEL
OF
AMPLIFIER
Meter multiplier and filament transformer (right) are mounted to the main panel. At left are
feedthrough capacitors for various supply leads. Disconnect plugs to auxiliary panel are at center.
Bandswitch shalt bracket is mounted to top of transformer.
The amplifier is attached to a dummy antenna or other r.f. load. The sideband exciter
may now be tuned and loaded, using the passive input resistor of the amplifier as a dummy
load. The filament of the 4CX1000A is turned
off, and high voltage applied to the amplifier.
The reading of the screen current meter is
noted, and the high voltage turned off and the
screen bleeder adjusted until the meter indicates 10 milliamperes of bleeder current.
The filament is now turned on, and the
plate voltage applied and checked. The "Adjust Bias" potentiometer at the rear of the amplifier is set to provide a static plate current
reading of 0.3 ampere. (Note that 60 ma. of
indicated meter reading is current drawn by
the screen regulator tubes and bleeder. This
current is constant, regardless of plate current,
and must be subtracted from the meter reading
to obtain true plate current.)
Next, apply a small carrier signal to the
amplifier to increase the plate current indication by about 50 ma. A large value of negative
screen current will be noted. Quickly set the
loading capacitor to full scale, and adjust the
plate tank capacitor for plate current dip,
which will be very small.
Monitor the screen current and advance the
grid drive until about plus 10 to 20 milliamperes of screen current are noted. Decrease
the capacitance of the loading capacitor (increase loading) slowly, and observe that the
screen current decreases as the loading in-
www.americanradiohistory.com
660
TH
H.F. Power Amplifiers
creases. Screen current will approach zero, and
perhaps go slightly negative. Re- resonate plate
tank, and increase excitation until plate current reaches 0.75 ampere. Screen current can
be adjusted by alternately varying grid drive
Figure 58
TOP VIEW OF AMPLIFIER CABINET
Placement of the major components may be
seen in this view. At left is 4CX1000A tube
and socket, with blower immediately behind
it. Atop the blower are the plate circuit ri.
choke and blocking capacitors. Ten meter section of tank coil is mounted in a vertical
position behind vacuum capacitor, which in
turn is mounted to tube enclosure. At right
is the tank inductor, with an auxiliary switch
deck (not used) mounted on rear of assembly.
This deck may be employed to switch antenna
relays. Lid of cabinet is perforated to provide
ventilation. Air intake is on left side of the
cabinet beside blower cage.
E
RADIO
and antenna loading. The sequence of events
is to tune, load, and vary the drive until a
plate meter reading of 0.75 ampere is achieved,
with an indicated screen current of approximately 0 to plus 20 ma. When excitation is
removed, screen current will drop to about
18 ma. This indicates a true resting screen
ma., plus a bleeder current of
current of
20 ma. Grid current, of course, is zero.
The carrier signal may now be removed, and
voice excitation applied to the amplifier. Plate
current may be "talked" up to about 0.38 ampere on voice peaks. True screen current will
run -5 ma. to plus 20 ma. on voice peaks,
depending upon the degree of loading and the
exact ratio between loading and grid drive.
Under optimum conditions, screen current will
-2
www.americanradiohistory.com
HANDBOOK
3 -1000Z
rest at 8 ma., and drop to about
voice waveforms.
29 -13
-2
ma. under
A 3 -1000Z
Linear Amplifier
The 3 -1000Z is a compact power triode intended to be used as a zero bias class B amplifier in audio or r.f. applications. Grounded
grid linear service is especially attractive, as
full legal input may be run at a plate potential
of only 2500 volts, yet the power gain of the
tube is high enough to allow sideband exciters
of the "100 watt" -type to drive it to full output. Neutralization of the grounded grid stage
is not necessary, as the excellent internal shielding of the tube reduces intra -stage feedback
to a minimum. Distortion products of this
amplifier are better than 35 decibels below
maximum p.e.p. level. A tuned cathode tank
circuit is employed in order to obtain greatest
linearity and power output.
Amplifier Circuit
The
3 -1000Z
amplifier
covers all amateur bands
between 3.5 Mc. and 29.7 Mc. with generous
overlaps. Bandswitching circuits are used, and
the unit is designed to operate with unbalanced
coaxial antenna systems having an s.w.r. of less
than 2/1. The complete schematic is given
661
in figure 61. A high -C, bandswitching cathode
circuit is used for best linearity (figure 62) .
The driving impedance of the 3 -1000Z is approximately 55 ohms, providing a close match
to the great majority of sideband exciters. The
"flywheel effect" of the cathode tank prevents
input waveform distortion caused by the halfcycle loading of the class B amplifier. Filament
voltage is fed to the tube via a shunt choke
(L2) placed in parallel with the tuned circuit.
The cathode coil is tapped for the various
amateur bands, and extra shunt capacity is
placed in the circuit to maintain the proper
C/L ratio at 3.5 Mc.
Plate current metering is accomplished in
the B -minus lead to the power supply to
remove dangerous voltages from the meter
movement. The meter is shunted with a wirewound resistor as a safety measure. For standby operation, the cathode to ground return of
the stage is opened by means of the voice
relay, and a small amount of idling current
flowing through a 50K cathode resistor provides sufficient bias to prevent annoying diode
noise from being generated during listening
periods. The voice relay shorts out the resistor
to allow normal operation of the stage.
It is necessary to "unground" the grounded
grid sufficiently to permit measurements of
grid current to be made. The 3 -1000Z has
ii
1(
Figure 59
NONINDUCTIVE
50 OHM GRID
Nine
Amplifier
RESISTOR
composition
2 -watt
resistors are immersed in
oil bath to provide high peak level of dissipation.
Exciter may be tuned up
using this resistor as a
dummy load, if desired.
www.americanradiohistory.com
H.F. Power Amplifiers
662
three grid pins, and each corresponding socket
terminal is bypassed to ground with a low
impedance r.f. shunt made of a 4.7 -ohm composition resistor and a 0.01 pfd., 1.2 kv.
ceramic disc capacitor connected in parallel.
Figure 60
2
KILOWATT P.E.P. GROUNDED -
GRID LINEAR AMPLIFIER
3 -10002 zero bias triode tube is
designed for grounded -grid linear amplifier
service, and is capable of full input at a
plate potential as low as 2500 volts. This
3 -1000Z amplifier covers all amateur bands
The Eimac
between 10 and 80 meters. Panel meters and
controls are (I. to r.): plate, Arid and output
meter; plate tuning (center); bandswitch;
cathode bandswitch and tuning (lower left);
antenna loading (center) and output voltmeter adjustment. Complete amplifier is enclosed in screen made of perforated Reynolds
aluminum sheet.
The voltage drop across the resulting resistance
(1.6 ohms) is measured by a simple d.c. voltmeter made up of a 0 -1 d.c. milliammeter
with a series multiplier chosen to provide a
full scale reading when 300 ma. of grid current develops 0.64 volts across the shunt. The
internal resistance of the meter is subtracted
from the value of the required series multiplier.
A pi- network tank circuit is used, with an
additional loading capacitor that can be
switched in the circuit to match low impedance
antenna loads commonly encountered on the
80 meter band. In addition, a diode voltmeter
is included to monitor the output voltage of
the amplifier.
The 3 -1000Z requires forced -air cooling to
maintain the base seals at a temperature below
200 °C, and the plate seal at a temperature
www.americanradiohistory.com
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RC -12130
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DAG
SR -50 MA. REPLACEMENT TYPE
SELENIUM RECTIFIER
IS
RESISTORS MARKED WITH ASTERISK K
PA /RS.
MAX.
ADJUST
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CLASS
MODULATOR
GRID CIRCUIT
UNLESS OTHERWISE INDICATED;
ALL RESISTORS 0.3 WATT
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Figure 13
SCHEMATIC, 15 WATT CLIPPER -AMPLIFIER
A three wire shielded cable should be used
to connect the 6B4 -G tubes to the driver transformer located at the grids of the class B
tubes. This cable may be any reasonable
length up to 25 or 30 feet. Any of the modulator configurations shown in figure 8 may be
driven with this simple speech amplifier.
30 -5
A 15 -Watt
Clipper - Amplifier
The near -ultimate in 'talk power" can be
obtained with low level clipping and filtering
combined with high level filtering. Such a
modulation system will have real "punch,"
yet will sound well rounded and normal. The
speech amplifier described in this section
makes use of low level clipping and filtering
and is specifically designed to drive a pair of
push -pull 810 modulators such as shown in
Section Three.
ping control, R2. Amplifier gain is controlled
by R,, in the grid circuit of the second section
of the 12AX7. A low pass filter having a 3500
cycle cut -off follows the 6AL5 clipper stage,
with an output of 5 volts peak audio signal
under maximum clipping conditions. A double triode 12AU7 cathode follower phase- inverter
follows the clipper stage and delivers a 125
volt r.m.s. signal to the push -pull grids of the
6B4 -G audio driver tubes. The 6B4 -G tubes
operate at a plate potential of 330 volts and
have a -68 volt bias voltage developed by a
small selenium rectifier supply applied to their
grid circuit. An audio output of 15 watts is developed across the secondary terminals of the
class B driver transformer with less than 5 per
cent distortion under conditions of no clipping.
A 5U4 -G and a choke input filter network provide unusually good voltage regulation of the
high voltage plate supply.
Circuit
Amplifier
Description
Construction
The schematic of the speech
amplifier -clipper is shown in
figure 13. A total of six tubes,
including a rectifier are employed and the unit
delivers 15 watts of heavily clipped audio.
A 12AX7 tube is used as a two stage microphone pre-amplifier and delivers approximately 20 volts (r.m.s.) audio signal to the
6AL5 series clipper tube. The clipping level
is adjustable between 0 db and 15 db by clip-
The clipper-amplifier may be
built upon the same chassis as
the power supply, provided
the low level stages of the amplifier are spaced
away from the power transformers and filter
chokes of the supply. All capacitors and resistors of the audio section should be mounted as
close to the respective sockets as is practical. For
minimum hum pickup, the filament leads to the
low level stages should be run in shielded braid.
www.americanradiohistory.com
HANDBOOK
15 -Watt
Those resistors in the 12AU7 phase inverter
plate circuit and the grid circuit of the 6B4 -G
tubes should be matched to achieve best phase
inverter balance. The exact value of the paired
resistors is not important, but care should be
taken that the values are equal. Random resistors may be matched on an ohmmeter to find
two units that are alike in value. When these
matched resistors are soldered in the circuit,
care should be taken that the heat of the soldering iron does not cause the resistors to
shift value. The resistors should be held firmly by the lead to be soldered with a long nose
pliers, which will act as a heat -sink between
the soldered joint and the body of the resistor.
If this precaution is taken the two phase inverter outputs will be in close balance.
When the wiring of the
speech amplifier has been
completed and checked, the
unit is ready to be tested.
Before the tubes are plugged in the amplifier,
the bias supply should be energized and the
voltage across the 600 ohm bleeder resistor
should be measured. It should be -68 volts.
Adjustment of
the Speech
Amplifier
If it is not, slight changes in the value of the
series resistor, Rr, should be made until the
correct voltage appears across the bleeder resistor. The tubes may now be inserted in the
amplifier and the positive and cathode voltages
checked in accordance with the measurements
given in figure 13. After the unit has been
Clipper -Amplifier
679
tested and is connected with the modulator,
R2 should be set so that it is impossible to over modulate the transmitter regardless of the setting of R1. The gain control (R.) may then be
adjusted to provide the desired level of clipping consistent with the setting of R2.
30 -6
A 200 -Watt
811 -A De -luxe Modulator
One of the most popular medium power r -f
amplifier stages consists of a single tetrode
tube, such as the 4 -125A, 813, or 7094 operating at a plate potential of 1200 - 1700 volts
and a plate current of 150 - 275 milliamperes.
Such an amplifier requires a minimum of r -f
driving power, allows an input of 300 to 400
watts, and yet employs power supply components that are relatively modest in cost. The
5 -db signal increase between a 300 watt transmitter and a 1000 watt transmitter is very
expensive when one considers the additional
cost of modulator and power supply equipment.
Additional economy may be achieved if
the modulator and final amplifier are operated
from the same power supply. The new series of
Chicago- Standard plate transformers provide
voltage ranges in the 1000 to 1500 volt region and are well suited for the combination
of this modulator and the aforementioned r -f
tubes. Within this voltage range, the 811 -A
triode is an excellent choice for the modulator
tubes. Zero bias operation may be had up to
Figure 14
REAR VIEW OF
811 -A MODULATOR
Modulator tubes and voltage
regulator are at right with
high level filter at center of
chassis. Plug -in speech amplifier is to left of clipper.
Gain and clipping controls
are atop the small chassis.
6L6'5881 is used as cathode
follower driver stage for
modulator.
www.americanradiohistory.com
THE RADIO
Speech and A. M. Equipment
680
PLUG -IN SPEECH AMPLIFIER
CRYSTAL
MIC
503
6AL5
12AX7
--
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12AU7
L1
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-ALL RESISTORS 1/Z
WATT UNLESS
0ÚF
OTHERWISE SPECIFIED.
-SW ITCH Si (PHONE-CW) SHOWN
IN PHONE POSITION.
1
6L6GB /5881
3- RY1 SHOWN ENERGIZED.
12
6
6L6GB
OF
ISO
10
AND
2D21 FILS
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6- 8#330 V TO EXCITER
7 -1
CW -PHONE SWITCH
6- I
9- GROUND, B1011-
RY1 CONTROL
12-MIC. CONTROL PUSH -TO-TALK CIRCUIT
7
e
811 -A
004
3NV
10
+HV OUr TO
AMPLIFIER
+HV IN
A
811
TS
RY1
2D21
Figure 15
SCHEMATIC, 200 WATT 811 -A MODULATOR
1, -1:3 Interstage Transformer. Stanco, A -53
"Poly -pedante" class B driver transformer. 2:1 ratio. Stanco, A -4761
T,-200 watt modulation transformer. 9 K primary. 5K secondary. Stanco, A -3829
T.-400 - 0 - 400 volts, 250 ma., 6.3 volts, 5 amperes. Stanco, PC -8413
T,-6.3 volts, 10 amperes. Stanco, P -6308
CH,-4 henry, 250 ma. Stanco, C -1412
L,-Low pass filter, 3000 cycle cut off. Chicago LPF -2.
"Splatter" filter, 300 ma. Stanco, C -2317
RY,-SPDT relay, high voltage insulation, 115 volt coil. Leach =1723 with 374 coils, or equivalent
Tr-
L,-
1250 volts, and only -4.5 volts is required
for 1500 volt operation. Bias voltage may be
obtained from flashlight batteries or other low
impedance source.
The 200 watt de -luxe modulator
is illustrated in figures 14 and
16 and the schematic is given
in figure 15. The low level audio stages are
similar to those of the speech amplifier shown
in Section Six. A 12AX7 is employed as two
Modulator
Circuit
stages of R -C amplification driving a 6AL5
speech clipper tube. A 3500 cycle low pass
filter follows the clipper, removing all high
order products of clipping action. A parallel connected 12AU7 follows the filter and is
transformer -coupled to a 5881 (6L6 -GB) cathode follower driver stage. The impedance of
the cathode circuit of the driver stage is extremely low; it provides an excellent driving
source for the class B modulator grid circuit.
www.americanradiohistory.com
HANDBOOK
81
Two 811 -A tubes are employed in the class
B stage. When operated at 1000 volts, no bias
supply is needed. At voltages of 1200 or above,
approximately 9 volts of bias is required. This
is supplied by a voltage divider composed of
a 20K, 10 watt resistor and a 2D21 thyratron
tube. When the miniature 2D21 is connected
as a triode, it acts as a voltage regulator tube
with a constant voltage drop of almost 9 volts
from plate to cathode. The tube will regulate
over 300 milliamperes of current while maintaining a reasonably constant voltage drop
across its terminals. The center tap of the
811 -A filament transformer (Ta) is thus held
at a positive potential with respect to ground.
Since the center tap of the 811 -A driver transformer (T2) is grounded, the modulator tubes
are biased at a constant negative voltage equal
to the voltage drop across the 2D21 regulator
tube in the cathode circuit of the class B stage.
The plate to plate load impedance of the
811 -A tubes when operating at 1500 volts is
approximately 12,000 ohms. A multi -match
type modulation transformer may be employed
if desired, but in this case a Stancor A -3829
unit was used. This transformer is designed to
match the plate -to -plate load impedance of
9,000 ohms to a secondary load of 5000 ohms.
With the 12,000 ohm load of the 811 -A tubes,
a secondary load of 7,500 ohms must be used
to maintain the same primary to secondary impedance ratio. This secondary load can be obtained with a single 7094 tube operating at
1500 volts and 200 milliamperes of plate current (300 watts input) . Other tubes and load
impedances can also be used, providing the
r -f input to the modulated stage does not exceed 400 watts. For example, a 4 -125A tube
operating at 2000 volts and 165 ma. (330
watts) may be modulated by this audio unit.
The secondary winding of the modulation
transformer can pass a maximum of 300 milliamperes with safety.
The audio output from the 811 -A stage is
passed through a high level low -pass "splatter
suppressor" which attenuates all audio frequencies above 3500 cycles. The use of both
low level and high level audio filters does
1
-A Modulator
681
much to reduce the broad sidebands and cochannel interference that seems to be so common on the amateur phone bands.
A high voltage relay RN', is employed to
short the secondary of the modulation transformer and remove plate potential from the
modulator tubes for c -w operation. The relay
is actuated by the "phone -c.w." switch on the
front panel of the modulator. Other segments
of this switch turn off the modulator filaments and provide extra contacts to control
auxiliary equipment.
A 350 volt supply is incorporated in the
modulator unit to power the speech amplifier
and driver stage and to provide power for
the r -f exciter stages of the transmitter. The
various power and control leads are brought
out to a multi -connector plug mounted on the
rear of the modulator deck.
Figure 16
UNDER -CHASSIS VIEW OF
811 -A MODULATOR
High voltage relay is between 811 -A tube
sockets, and low voltage components ore at
opposite end of chassis.
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682
Speech and A. M. Equipment
Modulator
Construction
The modulator is constructed
upon a steel chassis measur".A 101/2"
ing 8" x 17
aluminum panel is bolted to the chassis with
two mounting brackets to form a rugged assembly. Placement of the major parts may be
seen in figures 14 and 16. The modulation
transformer T3 and the 811 -A tubes occupy
the right end of the chassis, balanced in weight
by the power transformer T. and modulator
filament transformer T. at the opposite end of
the chassis. The center space is taken by the
plug -in speech amplifier, the high level splatter filter assembly and the 5881 driver stage.
The speech amplifier is constructed as a separate unit on a small aluminum utility box
measuring 5" x 3" x 2 ". The bottom of the
"x2
box holds two male plugs which match two
receptacles mounted on the amplifier chassis.
The speech amplifier, therefore, may be wired
and tested as a separate unit. Clipping and
audio level controls are mounted atop the
amplifier box as long usage of clipper circuits
has proven that these controls need not be readjusted once they are properly set.
The phone -c.w. switch, relay RY-1 and various small components are mounted beneath
the chassis (figure 16) . The input receptacle
for the speech amplifier box is located adjacent to the microphone receptacle on the front
panel of the modulator making the interconnecting lead less than two inches long. Also
placed beneath the chassis are the filter choke
for the low voltage supply and the various bypass and filter capacitors.
Wiring and Testing
the Modulator
The speech amplifier
should be wired first.
The small resistors and
capacitors are mounted either between the tube
socket pins, or between terminals of small
phenolic tie -point strips. Transformer T. is
fastened within the amplifier box and is wired
in the circuit after all other wiring is completed. Plugs PL, and PL: are mounted on the
bottom portion of the box; the plug pins are
wired to the proper points of the speech amplifier with short lengths of wire that allow
the bottom plate to be removed for inspection
and testing without the necessity of unsoldering any connections to the plugs.
The modulator chassis should be wired next.
All leads to T3, RY -3, and the low pass filter
should be carefully insulated from the chassis.
High voltage "5000 volt test" cable should be
employed for these connections. The capacitors that make up the high level audio filter
are mounted directly to the terminals of the
T
H
E
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filter choke which is mounted above the chassis on 1/2-inch ceramic insulators. High voltage
connections to the modulator are made through
Millen 37001 safety terminals.
When the wiring has been completed and
checked, the 12AX7, 6AL5, 12ÁU7, 5881,
and 5V4 -GB tubes should be inserted in their
sockets and the speech amplifier is plugged
into the modulator receptacles. The vertical
amplifier of an oscilloscope should be connected to one grid terminal of the 811 -A stage.
Plate voltage of the 5881 should be approximately 370 volts. A low level 1000 cycle tone
(approximately 0.05 volts, r.m.s.) is applied
to the amplifier input. The output level of the
speech amplifier is controlled by the setting
of the clipping control Rz and the audio gain
is controlled by potentiometer R1 in the grid
circuit of the 12AX7. The clipping control
should be set so that not more than 60 volts
r.m.s. is developed from one 811 -A grid to
ground. The modulator tubes may now be
plugged in their sockets. A 7K, 200 watt resistor should be placed between the "H.V.
Out" and " H.V. In" terminals, serving as a
dummy load, and 1500 volts applied to the
latter terminal. With no audio signal the
resting plate current of the modulator stage
should be approximately 15 milliamperes,
kicking up to about 160 milliamperes under
full output conditions. Final adjustment of the
clipping control may be made when the modulator is placed in use with the r -f section of
the transmitter. Potentiometer Rz is then adjusted to limit the peak modulation level under
sine wave modulation to approximately 90%.
V+
TO
DRi VER
STAGE
Figure 17
ZERO BIAS TETRODE MODULATOR
ELIMINATES SCREEN AND BIAS
SUPPLIES
driving power and simplicity are key
features of this novel modulator. Tubes ranging in size from 6AQ5's to 813's may be
employed in this circuit.
Low
T
-Class
8 driver transformer
Tr- Modulation
-6AQ5,
V,, V.
R,
R:
-Not
transformer
6L6, 807, 803, 813, etc.
used with 803 and 813
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HANDBOOK
683
803's
T
6 -WATT
SPEECH
AMP.
(F /G /2)
2
1
II
:
_
/70
DRIVING POWER=
TO
MODULATOR
LOAD
-
OPERATING CHARACTERISTICS
EGG (RMS)
ZS =6 25K
-
8+2500 V.
VOLTS
7 -6
WATTS
RESTING PLATE CUR.= SOMA.
MAX. PLATE CUR =340 MA.
POWER OUTPUT
=
5/0
SUPPRESSOR VOLT
=
WATTS
260 -340v
IIS
U
1
Figure 18
INEXPENSIVE 500 WATT
MODULATOR USING 803 TUBES
T,- "Poly -pedance"
Class B driver transformer 2:1 ratio. Stancor A -4761
18K primary,
6.25K secondary. Chicago CMS -3
Tr-500 watt output transformer.
T, -10 volts, 10 amperes.
M-0 - 500 mo.
Stancor
C -6461
Zero Bias
Tetrode Modulators
30 -7
Class B zero bias operation of tetrode tubes
is made possible by the application of the
driving signal to the two grids of the tubes as
shown in figure 17. Tubes such as the 6AQ5,
6L6, 807, 803, and 813 work well in this
circuit and neither a screen supply nor a bias
supply is required. The drive requirements
are low and the tubes operate with excellent
plate circuit efficiency. The series grid resistors for the small tubes are required to balance
the current drawn by the two grids, but are not
needed in the case of the 803 and 813 tubes.
Of great interest to the amateur is the circuit of figure 18, wherein 803 tubes are used
as high level modulators. These tubes will deliver 500 watts of audio in this configuration,
yet they require no screen or bias supply, and
can be driven by an 8 watt amplifier stage.
The use of 803 tubes (in contrast to 813's)
requires a higher level of driving power which
is offset by the fact that these tubes can often
be purchased "surplus" for less than four dollars. A pair of 6B4 tubes operating with cathode bias (figure 12) will suffice as a driving stage for the 803's. The power supply of
the speech amplifier provides high voltage for
the suppressors of the modulator stage.
Shown in figure 19 is a high level modulator using 813 tubes. A full 500 watts of
audio may be obtained at 2500 volts plate potential. Grid driving power is 5.5 watts. A
single 807 operating as a cathode follower
at 400 volts will provide sufficient drive for
the modulator stage. Plate to plate load impedance for the 813's is not critical. The Chicago CMS -3 500 watt modulation transformer
having a primary impedance of 18,000 ohms
has been used with success, although the optimum plate load impedance of the modulator
is closer to 20,000 ohms.
807
6C4,6J5 Erc.
T
813,5
TO
MODULATOR
LOAD
+400
DRIVER STAGE,OPERATING VOLTS,
MEASURED TO GROUND.
PIN 2
807
PIN
300
.3
PIN
26,5
NOTES
I
X
X
-EXACT VALUE OF 607 CATHODE RESISTOR
DEPENDS UPON RESISTANCE OF PRIMA RV
WINDING OF T2. ADJUST RESISTOR FOR
CATHODE 8 /AS OF 26.5 VOL rs, PLATE CURRENT OF 53 -55 MA.
2 -RMS OPERATING VOLTAGES AT MAXIMUM
OUTPUT SHOWN ON SCHEMATIC.
111
11SV.rt.
Figure 19
500 WATT MODULATOR USING 813 TUBES
T,-1:3 interstage transformer. Stancor A -53
Tr- "Poly -pedance" Class B driver transformer.
T,-500 watt output transformer. 18K primary,
T.-10 volts, 10 amperes, Stancor C-6461
2:1 ratio. Stancor A -4761
6.25K secondary. Chicago CMS -3
M-0
- 500 ma.
350 watts of audio are obtainable from this circuit at plate potential of 2000 volts.
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CHAPTER THIRTY -ON E
In view of the high cost of iron -core components such as go to make up the bulk of a
power supply, it is well to consider carefully
the design of a new or rebuilt transmitter in
terms of the minimum power supply requirements which will permit the desired performance to be obtained from the transmitter. Careful evaluation of the power supply requirements of alternative transmitter arrangements
will permit the selection of that transmitter
arrangement which requires the minimum of
power supply components, and which makes
most efficient use of such power supplies as
are required.
31 -1
Power Supply
Requirements
A power supply for a transmitter or for a
unit of station equipment should be designed
in such a manner that it is capable of delivering the required current at a specified voltage,
that it has a degree of regulation consistent
with the requirements of the application, that
its ripple level at full current is sufficiently
low for the load which will be fed, that its internal impedance is sufficiently low for the
job, and that none of the components shall be
overloaded with the type of operation contemplated.
The meeting of all the requirements of the
previous paragraph is not always a straightforward and simple problem. In many cases
compromises will be involved, particularly
when the power supply is for an amateur station and a number of components already on
hand must be fitted into the plan. As much
thought and planning should be devoted to the
power- supply complement of an amateur station as usually is allocated to the r -f and a -f
components of the station.
The arrival at the design for the power supply for use in a particular application may best
be accomplished through the use of a series
of steps, with reference to the data in this
chapter by determining the values of components to be used. The first step is to establish the operating requirements of the power
supply. In general these are:
1. Output voltage required under full load.
2. Minimum, normal, and peak output current.
3. Voltage regulation required over the current range.
684
www.americanradiohistory.com
Requirements
Figure
1
POWER SUPPLY
FILAMENT
CONTROL PANEL
VOLTAGE
PLATE
VOLTAGE
designed supply conhas separate primary
switches and indicator lamps
for the filament and plate circuits, overload circuit breaker,
plate voltage control switch
and primary circuit fuses.
A
685
well
trol panel
0
3500
411
PLATE VOLTAGE
FINAL
BIAS
2 AMP
2 AMP
SCREEN
2
AMP.
MN
4. Ripple voltage limit.
5. Rectifier circuit to be used.
The output voltage required of the power
supply is more or less established by the operating conditions of the tubes which it will supply. The current rating of the supply, however,
is not necessarily tied down by a particular
tube combination. It is always best to design
a power supply in such a manner that it will
have the greatest degree of flexibility; this
procedure will in many cases allow an existing power supply to be used without change
as a portion of a new transmitter or other item
of station equipment. So the current rating of
a new power supply should be established by
taking into consideration not only the requirements of the tubes which it immediately will
feed, but also with full consideration of the
best matching of power supply components in
the most economical current range which still
will meet the requirements. It is often long run economy, however, to allow for any likely
additional equipment to be added in the near
future.
Current- Rating
Considerations
The minimum current drain
which will be taken from a
power supply will be, in most
cases, merely the bleeder current. There are
many cases where a particular power supply
will always be used with a moderate or heavy
load upon it, but when the supply is a portion
of a transmitter it is best to consider the mini-
MOD
2AMMP.
EMISSION
FONECA
\
MN
mum drain as that of the bleeder. The minimum current drain from a power supply is of
importance since it, in conjunction with the
nominal voltage of the supply, determines the
minimum value of inductance which the input
choke must have to keep the voltage from soaring when the external load is removed.
The normal current rating of a power supply
usually is a round -number value chosen on the
basis of the transformers and chokes on hand
or available from the catalog of a reliable manufacturer. The current rating of a supply to
feed a steady load such as a receiver, a speech
amplifier, or a continuously-operating r -f stage
should be at least equal to the steady drain of
the load. However, other considerations come
into play in choosing the current rating for a
keyed amplifier, an amplifier of SSB signals,
or a class B modulator. In the case of a supply which will feed an intermittent load such
as these, the current ratings of the transformers and chokes may be lets than the maximum
current which will be taken; but the current
ratings of the rectifier tubes to be used should
be at least equal to the maximum current which
will be taken. That is to say that 300 -ma.
transformers and chokes may be used in the
supply for a modulator whose resting current
is 100 ma. but whose maximum current at
peak signal will rise to 500 ma. However, the
rectifier tubes should be capable of handling
the full 500 ma.
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686
T H
Power Supplies
The iron -core components of a power supply
which feeds an intermittent load may be chosen on the basis of the current as averaged
over a period of several minutes, since it is
heating effect of the current which is of greatest importance in establishing the ratings of
such components. Since iron -core components
have a relatively large amount of thermal inertia, the effect of an intermittent heavy current is offset to an extent by a key -up period
or a period of low modulation in the case of a
modulator. However, the current rating of a
rectifier tube is established by the magnitude
of the emission available from the filament of
the tube; the maximum emission must not be
exceeded even for a short period or the rectifier tube will be damaged. The above considerations are predicated, however, on the assumption that none of the iron -core components will
become saturated due to the high intermittent
current drain. If good -quality components of
generous weight are chosen, saturation will
not be encountered.
Voltage
Regulation
The general subject of voltage
regulation can really be divided
into two sub -problems, which differ greatly in degree. The first, and more
common, problem is the case of the normal
power supply for a transmitter modulator,
where the current drain from the supply may
vary over a ratio of four or five to one. In
this case we desire to keep the voltage change
under this varying load to a matter of 10 or
15 per cent of the operating voltage under full
load. This is a quite different problem from
the design of a power supply to deliver some
voltage in the vicinity of 250 volts to an oscillator which requires two or three milliamperes of plate current; but in this latter case
the voltage delivered to the oscillator must be
constant within a few volts with small variations in oscillator current and with large variations in the a -c line voltage which feeds the
oscillator power supply. An additional voltage
regulation problem, intermediate in degree between the other two, is the case where a load
must be fed with 10 to 100 watts of power at
a voltage below 500 volts, and still the voltage
variation with changes in load and changes
in a -c line voltage must be held to a few
volts at the output terminals.
These three problems are solved in the normal type of installation in quite different manners. The high -power case where output voltage must be held to within 10 to 15 per cent is
normally solved by using the proper value of
inductance for the input choke and proper
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value of bleeder at the output of the power
supply. The calculations are simple: the inductance of the power-supply input choke at
minimum current drain from the supply should
be equal in henries to the load resistance on
the supply (at minimum load current) divided
by 1000. This value of inductance is called
the critical inductance and it is the minimum
value of inductance which will keep the output voltage from soaring in a choke -input
power supply with minimum load upon the
output. The minimum load current may be
that due to the bleeder resistor alone, or it
may be due to the bleeder plus the minimum
drain of the modulator or amplifier to which
the supply is connected.
The low- voltage low- current supply, such as
would be used for a v.f.o. or the high- frequency oscillator in a receiver, usually is regulated with the aid of glow- discharge gaseous regulator tubes. These regulators are usually
called "VR tubes." Their use in various types
of power supplies is discussed in Section 31 -9.
The electronically -regulated power supply,
such as is used in the 10 to 100 watts power
output range, also is discussed later on in this
chapter,
The ripple -voltage limitation
imposed upon a power supply
is determined by the load
which will be fed by the supply. The tolerable
ripple voltage from a supply may vary from
perhaps 5 per cent for a class B or class C
amplifier which is to be used for a c -w stage
or amplifier of an FM signal down to a few
hundredths of one per cent for the plate-voltage supply to a low-level voltage amplifier in
a speech amplifier. The usual value of ripple
voltage which may be tolerated in the supply
for the majority of stages of a phone transmitter is between 0.1 and 2.0 per cent.
In general it may be stated that, with 60cycle line voltage and a single -phase rectifier
circuit, a power supply for the usual stages in
the amateur transmitter will be of the choke input type with a single pi- section filter following the input choke. A c -w amplifier or
other stage which will tolerate up to 5 per cent
ripple may be fed from a power supply whose
filter consists merely of an adequate -size input choke and a single filter capacitor.
A power supply with input choke and a
Ripple
Considerations
single capacitor also will serve in most cases
to feed a class B modulator, provided the output capacitor in the supply is sufficiently large.
The output capacitor in this case must be
capable of storing enough energy to supply the
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HANDBOOK
Requirements
peak- current requirement of the class B tubes
on modulation peaks. The output capacitor for
such a supply normally should be between 4
pfd. and 20 pfd.
Capacitances larger than 20 µfd. involve a
high initial charging current when the supply
is first turned on, so that an unusually large
input choke should be used ahead of the capacitor to limit the peak- current surge through
the rectifier tubes. A capacitance of less than
4 pfd. may reduce the power output capability
of a class B modulator when it is passing the
lower audio frequencies, and in addition may
superimpose a low -frequency "growl" on the
output signal. This growl will be apparent only
when the supply is delivering a relatively high
power output; it will not be present when modulation is at a low level.
When a stage such as a low -level audio amplifier requires an extremely low value of ripple voltage, but when regulation is not of importance to the operation of the stage, the high
degree of filtering usually is obtained through
the use of a resistance- capacitance filter.
This filter usually is employed in addition to
the choke -capacitor filter in the power supply
for the higher -level stages, but in some cases
when the supply is to be used only to feed
low- current stages the entire filter of the power supply will be of the resistance- capacitance
type. Design data for resistance -capacitance
filters is given in a following paragraph.
When a low- current stage requires very low
ripple in addition to excellent voltage regulation, the power supply filter often will end
with one or more gaseous -type voltage- regulator tubes. These VR tubes give a high degree
of filtering in addition to their voltage-regulating action, as is obvious from the fact that
the tubes tend to hold the voltage drop across
their elements to a very constant value regardless of the current passing through the tube.
The VR tube is quite satisfactory for improving both the regulation and ripple characteristics of a supply when the current drain will
not exceed 25 to 35 ma. depending upon the
type of VR tube. Some types are rated at a
maximum current drain of 30 ma. while others
are capable of passing up to 40 ma. without
damage. In any event the minimum current
through the VR tube will occur when the associated circuit is taking maximum current.
This minimum current requirement is 5 ma.
for all types of gaseous -type voltage -regulator
tubes.
Other types of voltage -regulation systems,
in addition to VR tubes, exhibit the added
RIPPLE IN TERMS OF C AT FULL LOAD
FULL- WAVE
RECTIFIER
TO
5 -25
NY
687
PERCENT RIPPLE
13.1
CAPACITANCE, C
2 UF
3 LF
25000
6.5
6.2
4.0
UF
6
L:
FIGURE 2
TO
RIPPLE IN TERMS OF LOAD RESISTANCE
FULL-WAVE
RECTIFIER
PERCENT RIPPLE
LOAD. ONMS
25000 (BLEEDER ONLY) 0.02
0.04
15000
25000
10000
0.06
5000
3000
2 000
0.1
O
17
0.25
FIGURE 3
TO FULL -WAVE
RIPPLE IN TERMS OF CI AND C2 AT FULL LOAD
RECTIFIER
C1
C2
2
2
1.2
0.7
3
25000
PERCENT RIPPLE
4
a
0
B
6
0.06
25
FIGURE 4
characteristic of offering a low value of ripple across their output terminals. The electronic -type of voltage -regulated power supply
is capable of delivering an extremely small
value of ripple across its output terminals,
even though the rectifier- filter system ahead
of the regulator delivers a relatively high
value of ripple, such as in the vicinity of 5 to
10 per cent. In fact, it is more or less self
evident that the better the regulation of such
a supply, the better will be its ripple characteristic. It must be remembered that the ripple
output of a voltage -regulated power supply of
any type will rise rapidly when the load upon
the supply is so high that the regulator begins
to lose control. This will occur in a supply of
the electronic type when the voltage ahead of
the series regulator tube falls below a value
equal to the sum of the minimum drop across
the tube at that value of current, plus the output voltage. In the case of a shunt regulator
of the VR -tube type, the regulating effect will
fail when the current through the VR tube
falls below the usual minimum value of about
5
ma.
Although figures 2, 3 and 4
give the value of ripple voltage for several more or less
standard types of filter systems, it is often of
value to be able to calculate the value of ripple voltage to be expected with a particular
set of filter components. Fortunately, the approximate ripple percentage for normal values
Calculation
of Ripple
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688
Power Supplies
e
Hr
T H
12 HT
FULL-WAVE
RECTIFIER
TO
Figure 5
SAMPLE FILTER FOR
CALCULATION OF RIPPLE
of filter components may be calculated with
the aid of rather simple formulas. In the two
formulas to follow it is assumed that the line
frequency is 60 cycles and that a full wave or
a full -wave bridge rectifier is being used. For
the case of a single -section choke -input filter
as illustrated in figure 2, or for the ripple at
the output of the first section of a two -section
choke input filter the equation is as follows,
118
Per cent ripple =
LC -1
where LC is the product of the input choke
inductance in henrys (at the operating current
to be used) and the capacitance which follows
this choke expressed in microfarads.
In the case of a two -section filter, the per
cent ripple at the output of the first section is
determined by the above formula. Then this
percentage is multiplied by the filter reduction
factor of the following section of filter. This
reduction factor is determined through the use
of the following formula:
LC -1
Filter reduction factor
1.76
Where LC again is the product of the inductance and capacitance of the filter section.
The reduction factor will turn out to be a decimal value, which is then multiplied by the percentage ripple obtained from the use of the
preceding formula.
As an example, take the case of the filter
diagrammed in figure 5. The LC product of the
first section is 16. So the ripple to be expected
at the output of the first section will be: 118/
(16 -1) or 118/15, which gives 7.87 per cent.
Then the second section, with an LC product
of 48, will give a reduction factor of: 1.76/
(48 -1) or 1.76/47 or 0.037. Then the ripple
percentage at the output of the total filter will
be: 7.87 times 0.037 or slightly greater than
0.29 per cent ripple.
-
Resistance-
Capacitance
Filters
In many applications where the
current drain is relatively small,
so that the voltage drop across
the series resistors would not be
E
R
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excessive, a filter system made up of resistors
and capacitors only may be used to advantage.
In the normal case, where the reactance of the
shunting capacitor is very much smaller than
the resistance of the load fed by the filter system, the ripple reduction per section is equal
to 1/ (2TrRC). In terms of the 120 -cycle ripple
from a full -wave rectifier the ripple -reduction
factor becomes: 1.33 /RC where R is expressed
in thousands of ohms and C in microfarads.
For 60 -cycle ripple the expression is: 2.66/RC
with R and C in the same quantities as above.
Filter System
Many persons have noticed,
particularly when using an input choke followed by a 2 -µfd.
first filter capacitor, that at some value of
load current the power supply will begin to
hum excessively and the rectifier tubes will
tend to flicker or one tube will seem to take
all the load while the other tube dims out. If
the power supply is shut off and then again
started, it may be the other tube which takes
the load; or first one tube and then the other
will take the load as the current drain is
varied. This condition, as well as other less
obvious phenomena such as a tendency for the
first filter capacitor to break down regardless
of its voltage rating or for rectifier tubes to
have short life, results from resonance in the
filter system following the high -voltage rectifier.
The condition of resonance is seldom encountered in low -voltage power supplies since
the capacitors used are usually high enough
so that resonance does not occur. But in high voltage power supplies, where both choke inductance and filter capacitance are more expensive, the condition of resonance happens
frequently. The product of inductance and capacitance which resonates at 120 cycles is
1.77. Thus a 1 -pfd. capacitor and a 1.77 henry
choke will resonate at 120 cycles. In almost
any normal case the LC product of any section
in the filter system will be somewhat greater
than 1.77, so that resonance at 120 cycles will
seldom take place. But the LC product for
resonance at 60 cycles is about 7.1. This is a
value frequently encountered in the input section of a high- voltage power supply. It occurs
with a 2 -pfd. capacitor and a choke which has
3.55 henrys of inductance at some current
value. With a 2-pfd. filter capacitor following
this choke, resonance will occur at the current
value which causes the inductance of the choke
to be 3.55 henrys. When this resonance does
occur, one rectifier tube (assuming mercuryResonance
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HANDBOOK
Rectification Circuits
689
vapor types) will dim and the other will become much brighter.
Thus we see that we must avoid the LC
products of 1.77 and 7.1. With a swinging -type
input choke, whose inductance varies over a
5 -to -1 range, we see that it is possible for
resonance to occur at 60 cycles at a low value
of current drain, and then for resonance to occur at 120 cycles at approximately full load
on the power supply. Since the LC product
must certainly be greater than 1.77 for satisfactory filtering along with peak- current limitation on the rectifier tubes, we see that with
a swinging -type input choke the LC product
must still be greater than 7.1 at maximum current drain from the power supply. To allow a
reasonable factor of safety, it will be well to
keep the LC product at maximum current drain
above the value 10.
It is possible to place the filter choke
in the B -minus lead of the power supply
which reduces the voltage potential appearing
from choke winding to ground. However, the
back e.m.f. of a good choke is quite high and
can develop a dangerous potential from the
center tap to ground on the secondary winding
of the plate transformer. If the transformer is
not designed to withstand this potential, it
is possible to break down the insulation at
this point.
31 -2
Rectification
Circuits
There are a large number of rectifier circuits
that may be used in the power supplies for station equipment. But the simpler circuits are
more satisfactory for the power levels up to
the maximum permitted the radio amateur. Figure 6 shows the three most common circuits
used in power supplies for amateur equipment.
Half -Wave
A half -wave rectifier, as shown in
Rectifiers
figure 6A, passes one half of the
wave of each cycle of the alternating current and blocks the other half. The
output current is of a pulsating nature, which
can be smoothed into pure, direct current by
means of filter circuits. Half -wave rectifiers
produce a pulsating current which has zero
output during one -half of each a -c cycle; this
makes it difficult to filter the output properly
into d.c. and also to secure good voltage regulation for varying loads.
Full -Wove
Rectifiers
A full-wave rectifier consists of
a pair of half -wave rectifiers
working on opposite halves of the
cycle, connected in such a manner that each
Figure 6
MOST COMMON RECTIFIER CIRCUITS
(A) shows a half -wove rectifier circuit, (8) is
the standard full -wove rectifier circuit used
with a dual rectifier or two rectifier tubes,
and (C) is the bridge rectifier circuit.
half of the rectified a -c wave is combined in
the output as shown in figure 7. This pulsating unidirectional current can be filtered to
any desired degree, depending upon the particular application for which the power supply
is designed.
A full -wave rectifier may consist of two
plates and a filament, either in a single glass
or metal envelope for low- voltage rectification
or in the form of two separate tubes, each having a single plate and filament for high -voltage
rectification. The plates are connected across
the high -voltage a -c power transformer winding, as shown in figure 6B. The power transformer is for the purpose of transforming the
110 -volt a -c line supply to the desired second-
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Power Supplies
TRANSFORMER
SECONDARY
VOLTAGE
RECTIFIED VOLTAGE
PLATE N1
RECTIFIED VOLTAGE
PLATE
2
N
COMBINED RECTIFIED
VOLTAGE
PLATES W1
0
l2
VOLTAGE
+
AFTER FIRST SECTION
OF FILTER
oL
D.C. VOLTAGE
AVAILABLE FOR
RADIO USE
Figure 7
FULL -WAVE RECTIFICATION
transformer secondary voltage, the
rectified output of each tube, the combined
output of the rectifier, the smoothed voltage
after one section of filter, and the substantially pure d.c. output of the rectifier- filter
after additional sections of filter.
Showing
ary a -c voltages for filament and plate supplies. The transformer delivers alternating current to the two plates of the rectifier tube; one
of these plates is positive at any instant during which the other is negative. The center
point of the high- voltage transformer winding
is usually grounded and is, therefore, at zero
voltage, thereby constituting the negative B
connection.
While one plate of the rectifier tube is conducting, the other is inoperative, and vice versa. The output voltages from the rectifier tubes
are connected together through the common
rectifier filament circuit. Thus the plates alternately supply pulsating current to the output (load) circuit. The rectifier tube filaments
or cathodes are always positive in polarity
with respect to the plate transformer in this
type of circuit.
The output current pulsates 120 times per
second for a full -wave rectifier connected to
a 60 -cycle a -c line supply; hence the output of
the rectifier must pass through a filter to
smooth the pulsations into direct current. Filters are designed to select or reject alternating
currents; those most commonly used in a -c
power supplies are of the low -past type.
Bridge
The bridge rectifier (figure 6C)
Rectification is a type of full -wave circuit
in which four rectifier elements
E
R
A D
I
O
or tubes are operated from a single high -voltage winding on the power transformer.
While twice as much output voltage can be
obtained from a bridge rectifier as from a center- tapped circuit, the permissible output current is only one-half as great for a given power
transformer. In the bridge circuit, four rectifiers and three filament heating transformer
windings are needed, as against two rectifiers
and one filament winding in the center- tapped
full -wave circuit. In a bridge rectifier circuit,
the inverse peak voltage impressed on any one
rectifier tube is halved, which means that tubes
of lower peak inverse voltage rating may be
used for a given voltage output.
Note that the center of the high voltage
winding of the bridge transformer (figure
6C) is not at ground potential. Many transformers having a center -tapped winding are
not designed for bridge service as the insulation between the center tap point and ground
is inadequate. Lack of insulation at this point
does no harm in a full -wave circuit, but may
cause breakdown when the transformer is used
in bridge configuration.
31 -3
Standard Power
Supply Circuits
Choke input is shown for all three of the
standard circuits of figure 6, since choke input
gives the best utilization of rectifier -tube and
power transformer capability, and in addition
gives much better regulation. Where greater
output voltage is a requirement, where the load
is relatively constant so that regulation is not
of great significance, and where the rectifier
tubes will be operated well within their peakcurrent ratings, the capacitor -input type of filter may be used.
The capacitor -input filter gives a no -load
output voltage equal approximately to the peak
voltage being applied to the rectifier tubes.
At full -load, the d -c output voltage is usually
slightly above one -half the secondary a -c voltage of the transformer, with the normal values
of capacitance at the input to the filter. With
large values of input capacitance, the output
voltage will run somewhat higher than the
r -m -s secondary voltage applied to the tubes,
but the peak current flowing through the rectifier tubes will be many times as great as the
d -c output current of the power supply. The
half -wave rectifier of figure 6A is commonly
used with capacitor input and resistance- capacitance filter as a high -voltage supply for a
cathode -ray tube. In this case the current drain
www.americanradiohistory.com
HANDBOOK
OA
Standard Circuits 691
HALF AND FULL VOLTAGE BRIDGE POWER SUPPLY
BO
TWO VOLTAGE BRIDGE POWER SUPPLY
T
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TWO TRANSFORMER POWER SUPPLY
OD CENTER TAPPED METHOD FOR UNTAPPED TRANSFORMERS
+
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TWO VOLTAGE POWER SUPPLY
A
FO
SPECIAL FILTER CIRCUIT FOR BRIDGE RECTIFIER
Figure 8
SPECIAL SINGLE PHASE RECTIFICATION CIRCUITS
description of the application and operation of each of these special circuits
accompanying text.
is very small so that the peak -current
rating
of the rectifier tube seldom will be exceeded.
The circuit of figure 6B is most commonly
used in medium -voltage power supplies since
this circuit is the most economical of filament
transformers, rectifier tubes and sockets, and
space. But the circuit of figure 6C, commonly
called the bridge rectifier, gives better transformer utilization so that the circuit is most
commonly used in higher powered supplies.
The circuit has the advantage that the entire
secondary of the transformer is in use at all
times, instead of each side being used alternately as in the case of the full -wave rectifier.
As a point of interest, the current flow through
the secondary of the plate transformer is a substantially pure a -c wave as a result of better
transformer utilization, instead of the pulsating d-c wave through each half of the power
transformer secondary in the case of the full wave rectifier.
The circuit of figure 6C will give the greatest value of output power for a given transformer weight and cost in a single -phase power
supply as illustrated. But in attempting to
bridge-rectify the whole secondary of a trans-
is
given
in
the
former designed for a full -wave rectifier, in
order to obtain doubled output voltage, make
sure that the insulation rating of the transformer to be used is adequate. In the bridge
rectifier circuit the center of the high -voltage
winding is at a d -c potential of one -half the
total voltage output from the rectifier. In a
normal full -wave rectifier the center of the
high -voltage winding is grounded. So in the
bridge rectifier the entire high -voltage secondary of the transformer is subjected to twice
the peak- voltage stress that would exist if the
same transformer were used in a full -wave rectifier. High -quality full -wave transformers will
withstand bridge operation quite satisfactorily
so long as the total output voltage from the
supply is less than perhaps 4500 volts. But
inexpensive transformers, whose insulation
is just sufficient for full -wave operation, will
break down when bridge rectification of the
entire secondary is attempted.
Special Single Phase Rectification
Circuits
www.americanradiohistory.com
Figure 8 shows six circuits which may prove
valuable when it is desired to obtain more than
692
T H
Power Supplies
E
R
A D
I
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PRIMARY
Eo
= 1.17
Es
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RIPPLE FREQUENCY= 3F
RIPPLE PERCENT = 18.3
PEAK INVERSE
2.09 Eo
TUBE VOLTAGE = = 2.44 Es
OA 3 -PHASE STAR
Figure 9
COMMON
Eo.
Is =
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1.35 Es
0.4os ID
.
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C.
RIPPLE FREQUENCY = 6F
RIPPLE PERCENT 4.2
PEAK INVERSE
2.09 Eo
TUBE VOLTAGE
2.53 Es
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6
-PHASE STAR
+Eo
Eo
PRIMARY
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=
RECTIFICATION
CIRCUITS
These circuits are used
when polyphase power is
available for the plate
supply of a high -power
transmitter. The circuit
at (B) is also called a
full -wave
three -phase
rectification system. The
circuits are described in
the accompanying text.
2.34 Es
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RIPPLE FREQUENCY =6F
RIPPLE PERCENT* 4.2
PEAK INVERSE
1.03 Eo
TUBE VOLTAGE .
2.44 Es
©
6
-PHASE BRIDGE
one output voltage from one plate transformer
or where some special combination of voltages
is required. Figure 8A shows a more or less
common method for obtaining full voltage and
half voltage from a bridge rectification circuit.
With this type of circuit separate input chokes
and filter systems are used on both output
voltages. If a transformer designed for use
with a full -wave rectifier is used in this circuit, the current drain from the full -voltage
tap is doubled and added to the drain from the
half-voltage tap to determine whether the rating of the transformer is being exceeded. Thus
if the transformer is rated at 1250 volts at 500
ma. it will be permissible to pull 250 ma. at
2500 volts with no drain from the 1250 -volt
tap, or the drain from the 1250 -volt tap may
be 200 ma. if the drain from the 2500 -volt
tap is 150 ma., and so forth.
Figure 813 shows a system which may be
convenient for obtaining two voltages which
are not in a ratio of 2 to 1 from a bridge -type
rectifier; a transformer with taps along the
winding is required for the circuit however.
With the circuit arrangement shown the voltage from the tap will be greater than one -half
the voltage at the top. If the circuit is changed
so that the plates of the two rectifier tubes
are connected to the outside of the winding
instead of to the taps, and the cathodes of the
other pair are connected to the taps instead
of to the outside, the total voltage ouput of
the rectifier will be the same, but the voltage
at the tap position will be lest than half the
top voltage.
An interesting variable- voltage circuit is
shown in figure 8C. The arrangement may be
used to increase or decrease the output voltage of a conventional power supply, as represented by transformer T,, by adding another
filament transformer to isolate the filament
circuits of the two rectifier tubes and adding
another plate transformer between the filaments
of the two tubes. The voltage contribution of
the added transformer T may be subtracted
from or added to the voltage produced by T1
www.americanradiohistory.com
HANDBOOK
Standard Circuits
simply by reversing the double -pole double throw switch S. A serious disadvantage of this
circuit is the fact that the entire secondary
winding of transformer T_ must be insulated
for the total output voltage of the power supply.
An arrangement for operating a full -wave
rectifier from a plate transformer not equipped
with a center tap is shown in figure 8D. The
two chokes L, must have high inductance ratings at the operating current of the plate supply to hold down the a -c current load on the
secondary of the transformer since the total
peak voltage output of the plate transformer
is impressed across the chokes alternately.
However, the chokes need only have half the
current rating of the filter choke L2 for a certain current drain from the power supply since
only half the current passes through each
choke. Also, the two chokes L, act as input
chokes so that an additional swinging choke
is not required for such a power supply.
A conventional two- voltage power supply
with grounded transformer center tap is shown
in figure 8E. The output voltages from this
circuit are separate and not additive as in
the circuit of figure 8B. Figure 8F is of advantage when it is desired to operate Class B
modulators from the half- voltage output of a
bridge power supply and the final amplifier
from the full voltage output. Both L, and L2
should be swinging chokes but the total drain
from the power supply passes through L, while
only the drain of the final amplifier passes
through 1.2. Capacitors C, and C2 need be rated
only half the maximum output voltage of the
power supply, plus the usual safety factor.
This arrangement is also of advantage in holding down the "key -up" voltage of a c -w transmitter since both L, and L2 are in series, and
their inductances are additive, insofar as the
"critical inductance" of a choke -input filter
is concerned. If 4 µfd. capacitors are used at
both C, and C2 adequate filter will be obtained
on both plate supplies for hum -free radiophone
operation.
Polyphase
It
Rectification
Circuits
cial
practice in commerinstallations
when the power drain from a
plate supply is to be greater
than about one kilowatt to use a polyphase rectification system. Such power supplies offer
better transformer utilization, less ripple output and better power factor in the load placed
upon the a -c line. However, such systems require a source of three -phase (or two -phase
with Scott connection) energy. Several of the
is usual
equipment
693
more common polyphase rectification circuits
with their significant characteristics are
shown in figure 9. The increase in ripple frequency and decrease in percentage of ripple
is apparent from the figures given in figure 9.
The circuit of figure 9C gives the best transformer utilization as does the bridge circuit
in the single -phase connection. The circuit
has the further advantage that there is no average d -c flow in the transformer, so that three
single -phase transformers may be used. A tap
at half -voltage may be taken at the junction
of the star transformers, but there will be d -c
flow in the transformer secondaries with the
power supply center tap in use. The circuit of
figure 9A has the disadvantage that there is an
average d -c flow in each of the windings.
Rectifiers Rectifying elements in high -voltage plate supplies are almost invariably electron tubes of either the high -vacuum or mercury-vapor type, although selenium
or silicon rectifier stacks containing a large
number of elements are often used. Low -voltage high -current supplies may use argon gas
rectifiers (Tungar tubes), selenium rectifiers,
or other types of dry -disc rectification elements. The xenon rectifier tubes offer some
advantage over mercury -vapor rectifiers for
high- voltage applications where extreme temperature ranges are likely to be encountered.
However, such rectifiers (3B25 for example)
are considerably more expensive than their
mercury -vapor counterparts.
Peak Inverse Plate In an a -c circuit, the maxi Voltage and Peak
mum peak voltage or cur Plate Current
rent is V 2 or 1.41 times
that indicated by the a -c
meters in the circuit. The meters read the root mean- square (r.m.s.) values, which are the
peak values divided by 1.41 for a sine wave.
If a potential of 1,000 r.m.s. volts is obtained from a high -voltage secondary winding
of a transformer, there will be 1,410 -volts peak
potential from the rectifier plate to ground. In
a single -phase supply the rectifier tube has
this voltage impressed on it, either positively
when the current flows or "inverse" when the
current is blocked on the other half -cycle. The
inverse peak voltage which the tube will stand
safely is used as a rating for rectifier tubes.
At higher voltages the tube is liable to arc
back, thereby destroying or damaging it. The
relations between peak inverse voltage, total
transformer voltage and filter output voltage
depend upon the characteristics of the filter
and rectifier circuits (whether full- or half wave, bridge, single -phase or polyphase, etc.).
www.americanradiohistory.com
694
T H
Power Supplies
=LINE VOLTS - HEATER VOLTS
HEATER AMPERES
+
Q
LINE RECTIFIER
C,..,
1-
CzT
SELENIUM
LINE RECTIFIER
1- o
VOLTAGE DOUBLER
FULL -WAVE
e
VOLTAGE DOUBLER
HALF -WAVE
SELENIUM
RECTIFIER
VOLTAGE
QUADRUPLER
Figure 10
TRANSFORMERLESS POWER-SUPPLY
CIRCUITS
Circuits such as shown above are also frequently called line -rectifier circuits. Selenium rectifiers, vacuum diodes, or gas diodes
may be used as the rectifying elements in
these circuits.
Rectifier tubes are also rated in terms of
peak plate current. The actual direct load current which can be drawn from a given rectifier
tube or tubes depends upon the type of filter
circuit. A full -wave rectifier with capacitor input passes a peak current several times the
direct load current.
In a filter with choke input, the peak current is not much greater than the load current
if the inductance of the choke is fairly high
(assuming full-wave rectification).
A full -wave rectifier with two rectifier elements requires a transformer which delivers
twice as much a -c voltage as would be the
case with a half -wave rectifier or bridge rectifier.
E
R
A D
I
O
Mercury -Vapor
Rectifier Tubes
The inexpensive mercury -vapor type of rectifier tube is
almost universally used in
the high- voltage plate supplies of amateur and
commercial transmitters. Most amateurs are
quite familiar with the use of these tubes but
it should be pointed out that when new or
long -unused mercury -vapor tubes are first
placed in service, the filaments should be operated at normal temperature for approximately twenty minutes before plate voltage
is applied, in order to remove all traces of
mercury from the cathode and to clear any
mercury deposits from the top of the envelope.
After this preliminary warm -up with a new
tube, plate voltage may be applied within 20
to 30 seconds after the time the filaments
are turned on, each time the power supply
is used. If plate voltage should be applied
before the filament is brought to full temperature, active material may be knocked from
the oxide-coated filament and the life of the
tube will be greatly shortened.
Small r -f chokes must sometimes be connected in series with the plate leads of mercury -vapor rectifier tubes in order to prevent
the generation of radio- frequency hash. These
r -f chokes must be wound with sufficiently
heavy wire to carry the load current and must
have enough inductance to attenuate the r -f
parasitic noise current to prevent it from flowing in the filter supply leads and then being
radiated into nearby receivers. Manufactured
mercury -vapor rectifier hash chokes are available in various current ratings from the James
Millen Company in Malden, Mass., and from
the J. W. Miller Company in Los Angeles.
When mercury -vapor rectifier tubes are operated in parallel in a power supply, small
resistors or small iron -core choke coils should
be connected in series with the plate lead of
each tube. These resistors or inductors tend
to create an equal division of plate current between parallel tubes and prevent one tube from
carrying the major portion of the current.
When high vacuum rectifiers are operated in
parallel, these chokes or resistors are not required.
Transformerless
Power Supplies
Figure 10 shows a group of
five different types of transformerless power supplies
which are operated directly from the a -c line.
Circuits of the general type are normally found
in a.c. -d.c. receivers but may be used in low powered exciters and in test instruments. When
circuits such as shown in (A) and (B) are
operated directly from the a -c line, the rec.
www.americanradiohistory.com
H A N D
B
O O K
tifier element simply rectifies the line voltage
and delivers the alternate half cycles of energy
to the filter network. With the normal type
of rectifier tube, load currents up to approximately 75 ma. may be employed. The d -c
voltage output of the filter will be slightly
less than the r -m -s line voltage, depending
upon the particular type of rectifier tube employed. With the introduction of the miniature
selenium rectifier, the transformerless power
supply has become a very convenient source
of moderate voltage at currents up to perhaps
500 ma. A number of advantages are offered
by the selenium rectifier as compared to the
vacuum tube rectifier. Outstanding among
these are the factors that the selenium rectifier operates instantly, and that it requires no
heater power in order to obtain emission. The
amount of heat developed by the selenium rectifier is very much less than that produced by
an equivalent vacuum -tube type of rectifier.
In the circuits of figure 10 (A), (B) and
(C) , capacitors G and G should be rated
at approximately 150 volts and for a normal
degree of filtering and capacitance, should be
between 15 to 60 ,dd. In the circuit of figure
10D, capacitor C, should be rated at 150
volts and capacitor G should be rated at 300
volts. In the circuit of figure 10E, capacitors
C, and G should be rated at 150 volts and
G and G should be rated at 300 volts.
The d -c output voltage of the line rectifier
may be stabilized by means of a VR tube.
However, due to the unusually low internal
resistance of the selenium rectifier, transform erless power supplies using this type of rectifying element can normally be expected to
give very good regulation.
Voltage -Doubler Figures IOC and 1OD illusCircuits
trate two simple voltage-
doubler circuits which will
deliver a d -c output voltage equal approximately to twice the r -m -s value of the power line
voltage. The no -load d -c output voltage is
equal to 2.82 times the r -m -s line voltage
value. At high current levels, the output voltage will be slightly under twice the line voltage. The circuit of figure IOC is of advantage
when the lowest level of ripple is required
from the power supply, since its ripple frequency is equal to twice the line frequency.
The circuit of figure 10D is of advantage when
it is desired to use the grounded side of the
a -c line in a permanent installation as the return circuit for the power supply. However,
with the circuit of figure IOD the ripple frequency is the same as the a -c line frequency.
Standard Circuits
695
OUTSIDE COLLECTOR
INSIDE COLLECTOR
PHENOLIC WASNEF
BASE PLATE
SELENIUM COAT
100
90
-
SELENIUM RECTIFIER CELL
60
Lß
>
70
60
U 50
W 40
U 30
20
W
00
C,
50
100
150
200
250
300
RELATIVE LOAD CURRENT,
PERCENT' OF FULL LOAD
Figure 11
THE SELENIUM RECTIFIER
A -The selenium rectifier is a semi -conductor
stack built up of nickel plated aluminum
discs
coated on one side
with selenium
Rectifier efficiency
8- alloy.
high, reaching 70";
for single phase service, dropping slightly
is
at high current densities.
Voltage
The circuit of figure 10E illustrates a voltage quadrupler circuit for miniature selenium rectifiers. In effect this circuit is equivalent to
two voltage doublers of the type shown in figure 10D with their outputs connected in series.
The circuit delivers a d -c output voltage under
light load approximately equal to four times
the r -m -s value of the line voltage. The noload d -c output voltage delivered by the quadrupler is equal to 5.66 times the r -m -s line
voltage value and the output voltage decreases
rather rapidly as the load current is increased.
In each of the circuits in figure 10 where
selenium rectifiers have been shown, conventional high -vacuum rectifiers may be substituted with their filaments connected in series
and an appropriate value of the line resistor
added in series with the filament string.
Quadrupler
31
-4
Selenium and
Silicon Rectifiers
Selenium rectifiers are characterized by long
life, dependability, and maintenance -free operation under severe operating conditions. The
THE RADIO
Power Supplies
696
POSITIVE TERMINAL
NEGATIVE TERMINAL
CONTACT
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Irrror
11711.
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SILICON CELL
50
O
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¶50
LOAD CURRENT, PEPCEN7
250
200
OF FULL LOAD
300
Figure 12
VOLTAGE REGULATION OF
SELENIUM CELL
This graph applies to single phase lull wove
bridge, and center -tap circuits which utilize
both halves of the input wave. In single phase
hall wave circuits the regulation will be poorer.
selenium rectifier consists of a nickel -plated
aluminum base plate coated with selenium
over which a low temperature alloy is sprayed.
The base plate serves as the negative electrode
and the alloy as the positive, with current
flowing readily from the base plate to the
alloy but encountering high resistance in the
opposite direction (figure 11A). This action
results in effective rectification of an alternating input voltage and current with the efficiency of conversion dependent to some extent
upon the ratio of the resistance in the conducting direction to that of the blocking direction. In normal power applications a ratio
of 100 to 1 is satisfactory; however, special
applications such as magnetic amplifiers often
require ratios in the order of 1000 to 1.
The basic selenium rectifier cell is actually
a diode capable of half wave rectification.
Since many applications require full wave rectification for maximum efficiency and minimum ripple, a plurality of cells in series, parallel, or series -parallel combinations are stacked
in an assembly.
Selenium rectifiers are operated over a wide
range of voltages and currents. Typical applications range from a few volts at milliamperes
of current to thousands of amperes at relatively high voltages.
The efficiency of high quality selenium rectifiers is high, usually in the order of 90%
in three phase bridge circuits and 70% in
single phase bridge circuits. Of particular interest is the very slight decrease in efficiency
even at high current overloads (figure 11B).
Threshold Voltage
and Aging
A minimum voltage is required to permit a selen-
ium rectifier to conduct
in the forward direction. This voltage, commonly known as the threshold voltage, precludes the use of selenium rectifiers at ex-
SPRING
Figure 13
THE SILICON CELL
The common silicon rectifier is
a
pressure
contact device capable of operation in ambient temperatures as high as 150 °C. Heavy
end ferrules that fit standard fuse clips are
large enough to provide "heat sink" action.
The positive ferrule is grooved to provide
polarity identification and prevent incorrect
mounting.
tremely low ( less than one volt) applications.
The threshold voltage will vary with temperature and will increase with a decrease in
temperature.
Under operating conditions, and to a lesser
extent when idle, the selenium rectifier will
age. During the aging period the forward
resistance will gradually increase, stabilizing
at a new, higher value after about one year.
This aging will result in approximately a 7%
decrease in output voltage.
Voltage
Regulation
The selenium rectifier has extremely low internal impedance
which exhibits non -linear characteristics with respect to applied voltage. This
results in good voltage regulation even at large
overload currents. Figure 12 shows that as the
load is varied from zero to 300% of normal,
the output voltage will change about 10 %.
It should be noted that because of non -linear
characteristics, the voltage drop increases rapidly below 50% of normal load.
Of all recent developments in the
field of semi -conductors, silicon
rectifiers offer the most promising
range of applications; from extreme cold to
high temperature, and from a few watts of
output power to very high voltage and currents. Inherent characteristics of silicon allow
junction temperatures in the order of 200 °C
before the material exhibits intrinsic properties. This extends the operating range of silicon devices beyond that of any other efficient
semi -conductor and the excellent thermal range
coupled with very small size per watt of output power make silicon rectifiers applicable
where other rectifiers were previously considered impractical.
Silicon
Rectifiers
Silicon
Current Density
www.americanradiohistory.com
The current density of a silicon rectifier is very high, and
on present designs ranges
HANDBOOK
Mobile Power Supply
697
from 600 to 900 amperes per square inch of
effective barrier layer. The usable current density depends upon the general construction of
the unit and the ability of the heat sink to
conduct heat from the crystal. The small size
of the crystal is illustrated by the fact that a
rectifier rated at 15 amperes d.c., and 150
amperes peak surge current has a total cell
volume of only .00023 inches. Peak currents
are extremely critical because the small mass
of the cell will heat instantaneously and could
reach failure temperatures within a time lapse
of microseconds. The assembly of a typical
silicon cell is shown in figure 13.
The reverse direction of a silicon rectifier is characterized
by extremely high resistance,
up to 10" ohms below a critical voltage point.
This point of avalanche voltage is the region
of a sharp break in the resistance curve, followed by rapidly decreasing resistance (figure
15A). In practice, the peak inverse working
voltage is usually set at least 20% below the
avalanche point to provide a safety factor.
The forward direction, or direction of low
resistance determines the majority of power
loss within the semi -conductor device. Figure
15B shows the static forward current characteristics versus applied voltage. The threshold
voltage is about 0.6 volts.
Since the forward resistance of a semi-conductor is very low, any unbalance between
threshold voltages or internal voltage drop
would cause serious unbalance of load distribution and ultimate failure of the overloaded
section. A small resistance should therefore
be placed in series with each half wave section
Operating
Characteristics
Figure 14
MINIATURE SEMI- CONDUCTOR
TYPE RECTIFIER
Raytheon CK -777 power rectifier bolts to
chassis to gain large "heat sink" area. Low
internal voltage drop and high efficiency
permit small size of unit.
operating in parallel to balance the load currents.
Some interesting and practical semi- conductor power supplies are shown in figure 16.
Remember that the circuits of figure 16A and
B, and those of figure 10 are "hot" with respect to one side of the power line.
31 -5
100 Watt Mobile
Power Supply
High efficiency and compact size are the
most important factors in the design of mobile
power supplies. The power package described
in this section meets these stringent require-
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.t
.t..
,
.
-50
-60
..ri,ulsu/.a-
-70
-80
o
MI M1111111111110
=
i=M.iiTT.TÜ,INIGi'yf.
ti=t...Rrlsa
-
1.
-
ti
19 -70
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t.t...ttl-twncr.trn
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9
1
2
POWER
r are ht
-
IN
I11
_GEE -80
MO
3
4
-90
6
Bard an .eei reW
Figure 9.
CONVERSION CHART: POWER TO DECIBELS
Power levels between 6 micromicrowatts and 6000 watts may be referred to corresponding decibel
levels between -90 and 60 db, and vice versa, by means of the above chart. Fifteen ranges are
provided. Each curve begins at the same point where the preceding one ends, enabling uninterrupted
go of the wide db and power ranges with condensed chart. For example: the lowermost curve
ends at -00 db or 60 micromlcrowatts and the next range starts at the same level. Zeno db level is
taken as 6 milliwatts (.006 watt).
www.americanradiohistory.com
J
W
Il
sltaer:
I.t...
-20
MI
INIMMIMI..ll.
Rtta..
t.MM..11i\+It61=
itit.t...m
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773
It is often convenient
to he able to convert a
decibel value to a power equivalent. The formula used for this operConverting Decibels
to Power
ation
THE RADIO
Radio Mathematics and Calculations
774
Solution:
--17.3
2.7
- 20
is
P
=
Nan
10
0.006 X antilog N'ö
where P is the desired level in watts and Nab
the decibels to be converted.
To determine the power level P from a decibel equivalent, simply divide the decibel value
by 10; then take the number comprising the
antilog and multiply it by 0.006; the product
gives the level in watts.
Note: In problems dealing with the conversion of miner decibels to power, it often happens that the decibel value -Na, is not
divisible by 10. When this is the case,
the numerator in the factor
-
-
N,,n
10
--
must be
made evenly divisible by 10, the negative signs
must be observed, and the quotient labeled accordingly.
To make the numerator evenly divisible by
10 proceed as follows: Assume, for example,
is some such value as -38; to
that
make this figure evenly divisible by 10, we
must add -2 to it, and, since we have added
a negative 2 to it, we must also add a positive
2 so as to keep the net result the same.
Our decibel value now stands, -40 + 2.
Dividing both of these figures by 10, as in the
4 and +0.2. Putequation above, we have
ting the two together we have the logarithm
-4.2 with the negative characteristic and the
positive mantissa as required.
The following examples will show the technique to be followed in practical problems.
(a) The output of a certain device is rated
at -74 db. What is the power equivalent?
Solution:
-
N,ib
-
-74 not evenly divisible
NÖ,
(
Routine:
-- 80
74
+6
6
Nni
10
-
by 10)
+6
-80 +
10
6
-
8.6
ontilog -8.6 = 0.000 000 04
.006 X 0.000 000 04 =
0.000 000 000 24 watt or
240 micro- microwatt
(b) This example differs somewhat from
that of the foregoing one in that the mantissas
are added differently. A low- powered amplifier
has an input signal level of -17.3 db. How
many milliwatts does this value represent?
-
-20 +
Antilog
+
2.7
2.7
+ 2.7
-2.27
10
-2.27 =
0.0186
0.006 X 0.0186 = 0.000
0.1116 milliwatt
1
1
16
watt or
Input voltages: To determine the required
input voltage, take the peak voltage necessary
to drive the last class A amplifier tube to maximum output, and divide this figure by the total overall voltage gain of the preceding stages.
Computing Specifications: From the preceding explanations the following data can be
computed with any degree of accuracy warranted by the circumstances:
(1) Voltage amplification
(2) Overall gain in db
(3) Output signal level in db
(4) Input signal level in db
(5) Input signal level in watts
(6) Input signal voltage
When a power level is available which must
be brought up to a new power level, the gain
required in the intervening amplifier is equal
to the difference between the two levels in decibels. If the required input of an amplifier for
full output is -30 decibels and the output
from a device to be used is but -45 decibels,
the pre -amplifier required should have a gain
of the difference, or 15 decibels. Again this is
true only if the two amplifiers are properly
matched and no losses are introduced due to
mismatching.
Push -Pull
Amplifiers
To double the output of any cas cade amplifier, it is only necessary to connect in push -pull the
last amplifying stage, and replace the inter stage and output transformers with push -pull
types.
To determine the voltage gain (voltage ratio) of a push -pull amplifier, take the ratio of
one hall of the secondary winding of the push.
pull transformer and multiply it by the
of
one of the output tubes in the push -pull stage;
the product, when doubled, will be the voltage
amplification, or step-up.
a
Other Units and
Zero Levels
When working with deci-
bels one should not immediately take for granted
that the zero level is 6 milliwatts for there are
other zero levels in use.
In broadcast stations an entirely new system
is now employed. Measurements made in
www.americanradiohistory.com
HANDBOOK
Trigonometry
acoustics are now made with the standard zero
level of 10-'° watts per square cm.
Microphones are often rated with reference
to the following zero level: one volt at open
circuit u hen the sound pressure is one millibar.
In any case, the rating of the microphone must
include the loudness of the sound. It is obvious
that this zero level does not lend itself readily
for the calculation of required gain in an am-
plifier.
The VU: So far, the decibel has always referred to a type of signal which can readily be
measured, that is, a steady signal of a single
frequency. But what would be the power level
of a signal which is constantly varying in volume and frequency? The measurement of voltage would depend on the type of instrument
employed, whether it is measured with a
thermal square law meter or one that shows
average values; also, the inertia of the movement will change its indications at the peaks
and valleys.
After considerable consultation, the broadcast chains and the Bell System have agreed
on the VU. The level in VU is the level in
milltu att zero level and measdecibels above
ured with a carefully defined type of instrument across a 600 ohm line. So long as we
deal with an unvarying sound, the level in VU
milliwatt; but
is equal to decibels above
when the sound level varies, the unit is the
VU and the special meter must be used. There
is then no equivalent in decibels.
The Neper: We might have used the natural
logarithm instead of the common logarithm
when defining our logarithmic unit of sound.
This was done in Europe and the unit obtained
is known as the neper or napier. It is still
found in some American literature on filters.
1
1
1
1
neper = 8.686 decibels
decibel = 0.1151 neper
AC Meters With
Decibel Scales
test instruments
are now equipped with
scales calibrated in deci-
Many
bels which is very handy when making measurements of frequency characteristics and gain.
These meters are generally calibrated for connection across a 500 ohm line and for a zero
level of 6 milliwatts. When they are connected
across another impedance, the reading on the
meter is no longer correct for the zero level of
6 milliwatts. A correction factor should be
applied consisting in the addition or subtraction of a steady figure to all readings on the
meter. This figure is given by the equation:
db to be added
=
10 log
szo
where Z is the impedance of the circuit under
measurement.
SECOND
FIRST
QUADRANT
QUADRANT
THIRD
FOURTH
QUADRANT
QUADRANT
775
Figure 10.
CIRCLE IS DIVIDED INTO
THE
FOUR QUADRANTS BY TWO PER-
PENDICULAR LINES AT RIGHT
ANGLES TO EACH OTHER.
"northeast" quadrant thus formed is
known os the first quadrant; the others are
The
numbered consecutively in a counterclockwise
direction.
Trigonometry
Trigonometry is the science of
mensuration of triangles. At first
and Use
glance triangles may seem to
have little to do with electrical phenomena;
however, in a.c. work most currents and voltages follow laws equivalent to those of the
various trigonometric relations which we are
about to examine briefly. Examples of their
application to a.c. work will he given in the
Definition
section on Vectors.
Angles are measured in degiees or in radians. The circle has been divided into 360
degrees, each degree into 60 minutes, and each
minute into 60 seconds. A decimal division of
the degree is also in use because it makes calculation easier. Degrees, minutes and seconds
are indicated by the following signs: °, ' and "
Example: 6° 5' 23" means six degrees, five
minutes, twenty -three seconds. In the decimal
notation we simply write 8.47 °, eight and
forty -seven hundredths of a degree.
When
a
circle is divided into four quadrants
by two perpendicular lines passing through
the center (Figure 10) the angle made by the
two lines is 90 degrees, known as a right angle.
Two right angles, or 180° equals a straight
angle.
The radian: If we take the radius of a circle
and bend it so it can cover a part of the circumference, the arc it covers subtends an angle
called a radian (Figure 11 ). Since the diameter, of a circle equals 2 times the radius,
there are 27 radians in 360 °. So we have the
following relations:
7= 3.14159
1 radian =57° 17'45 " = 57.2958°
1 degree = 0.01745 radians
/2 radians =90°
yr radians =180 °'
7/3 radians =60°
www.americanradiohistory.com
776
THE RADIO
Radio Mathematics and Calculations
In the angle A, Figure 13A, a line is drawn
from P, perpendicular to b. Regardless of the
point selected for P, the ratio a/c will always
be the same for any given angle, A. So will all
the other proportions between a, b, and c remain constant regardless of the position of
point P on c. The six possible ratios each are
named and defined as follows:
a
sine A =
Figure 11.
THE RADIAN.
tangent
radian is an angle whose arc is exactly equal
to the length of either side. Note that the
angle is constant regardless of the length of
the side and the arc so long as they are equal.
A radian equals 57.2958
A
a
=
A
.
In trigonometry we consider an angle generated by two lines, one stationary and the
other rotating as if it were hinged at 0, Figure
12. Angles can be greater than 180 degrees and
even greater than 360 degrees as illustrated in
this figure.
Two angles are complements of each other
when their sumis 90 °, or a right angle. A is
the complement of B and B is the complement
of A when
A=(90° -B)
cosine A
c
cosecant A =
sin. 60°
_
a
t/2 1(3
=
a
and
a
1/2
=bc
=
1/2
1/2
1/2
cot 60°
-
=
1
=
1/2
sec 60°
= b
1
=
1/2
=2
csc 60°
=
a
1
=
IS
2/j
1/2
Another example: Let the angle be 45 °, then
the relations between the lengths of a, b, and
c are as shown in Figure 13C, and the six
functions are:
AC
AN ANGLE
=1/2
1/2 V-3
tan 60° = b =
Two angles are supplements of each other
when their sum is equal to'a straight angle, or
180 °. A is the supplement of B and B is the
supplement of A when
B= (180 ° -A)
=
cos 60 °
-A)
A= (180 ° -B)
c
Let us take a special angle as an example.
For instance, let the angle A be 60 degrees as
in Figure 13B. Then the relations between the
sides are as in the figure and the six functions
become:
and when
B= (90°
b
cotangent A =
C
secant A =
=
Figure 12.
GENERATED BY TWO LINES, ONE STATIONARY AND THE OTHER
ROTATING.
stationary; the line with the small arrow at the far end rotates in a counterclockwise
direction. At the position illustrated in the lefthandmost section of the drawing it makes an angle,
A, which is less thon 90 and is therefore in the first quadrant. In the position shown in the second
portion of the drawing the angle A has increased to such a value that it now lies in the third
quadrant; note that an angle can be greater than 180 . In the third illustration the angle A is in
the fourth quadrant. In the fourth position the rotating vector has made more than one complete
revolution and is hence in the fifth quadrant; since the fifth quadrant is an exact repetition of the
first quadrant, its values will be the same os in the lefthandmost aortion of the illustration.
The line OX is
www.americanradiohistory.com
Trigonometric Relations
H A N D B O O K
777
P
C=e
90"
O
b=0
Figure 13.
THE TRIGONOMETRIC FUNCTIONS.
In the right triangle shown
b and c; the trigonometric
sides a, b and c. In (B) are
is 1. In (C) angle A is IS ;
sin 45°
in (A) the side opposite the ongle A is o, while the adjoining sides are
functions of the ongle A are completely defined by the ratios of the
shown the lengths of the sides a and b when angle A is 60- and side c
a and b equal 1, while c equals V-2. In (D) note that c equals a for a
right angle while b equals O.
=
1/, V-2
tan 45° =
1
= 1z Ari
1 =
cot 45° =
1
=
sec 45°
°= --=
sec
a
=
b
-
a
0
cos
12
_
cos A
c
c
co
b
When the angle
values are then:
sin 0
sec 0°
a
c
=
c
b
c
=
1
c
0
a
-
o
=
o
C
_ -a =
cosec 90°
is zero, a
-= tan 0° =
b0
A
-cot
A
=0
=sin (complement of A) =sin (90 ° -A)
the right triangle of Figure 15,
A =b /c =sin B and B=90° -A or the
complement of A. For the same reason:
because in
cos
cotA = tan (90 ° -A)
csc
A =
=0
and b
=
0
cos 0°
0
cot 0° =
b
=c
identities:
tan A = a/b
cot A = b/a
=
=
a
b
b
a
tan A
cot A
1
CO
In the same triangle we can do the same for
functions of the angle B
0
a
cosec0° =
sin A
For the same reason we have the following
1
=c. The
=
c
-A)
(90 °
sec
In the right triangle of
Figure 15, sin A =a /c and
by transposition
Relations in
Right Triangles
a
-=-=
90°
b
90° =
cot 90° =
=
1
sec
From the definitions also follows the relation
a
1
A -
cos
and tan A
1
There are some special difficulties when the
angle is zero or 90 degrees. In Figure 13D an
angle of 90 degrees is shown; drawing a line
perpendicular to b from point P makes it fall
on top of c. Therefore in this case a = c and
b = 0. The six ratios are now:
tan 90°
I
cosec A
1
V-2-
cosec 45°
sin 90
sin A =
1
=
1
-
_
cos 45°
It follows from the definitions that
Relations Between
Functions
1
_
c
a
_c
0-
co
In general, for every angle, there will be definite values of the six functions. Conversely,
when any of the six functions is known, the
angle is defined. Tables have been calculated
giving the value of the functions for angles.
From the foregoing we can make up a small
table of our own (Figure 14), giving values of
the functions for some common angles.
Angle
Sin
Cos.
Tan
Cot
Sec.
0
0
1
0
m
1
30°
45°
1/2
60°
1/2
90°
1/2
1/2
\
13
1
1/2
0
1/3
\
VT
3 1/3 V-3-
1
1/2
1
1/3
w
0
v,
\ri
2/3
2
Cosec.
2
V
2/3
cc
Figure 14.
Values of trigonometric functions for common
angles in the first quadrant.
www.americanradiohistory.com
'
VT
1
THE RADIO
Radio Mathematics and Calculations
778
POSITIVE FUNCTIONS
QUADRANT
FIRST
SECOND QUADRANT
sins, cosec
all functions
b
Figure 15.
in this figure the sides a, b, and c are used
to define the trigonometric functions of angle
B as well as angle A.
sin B = b/c
cos B = a/c
tan B = b/a
b=csin
cot B
a
= a/b
Functions of Angles
Greater than
90 Degrees
a
b
=
=
=
B
tan, cot
b cot B
In angles greater than
90 degrees, the values
of a and b become negative on occasion in ac-
THIRD QUADRANT
=
tan A
=
sec
a
= pos.
cos
-b
= neg.
cot A
A = -c
neg.
c
a
cosec
A =
=
A =
-b
-
c
a
=
tan A =
b-a
-a
=
neg.
cos
=
pos.
cot A =
A =
sec
sin A
-b
= neg.
= pos.
=
tan A =
= pos.
-b
A =
= neg.
cosec A
=
-
-c a
= neg.
And in the fourth quadrant (270° to 360 °):
neg.
For an angle in the third quadrant (180° to
270 °), the functions are
sin A
FUNCTIONS.
The functions listed in this diagram are positive; all other functions are negative.
neg.
--ab
c
=
FOURTH QUADRANT
Figure 17.
SIGNS OF THE TRIGONOMETRIC
cordance with the rules of Cartesian coordinates. When b is measured from 0 towards
the left it is considered negative and similarly,
when a is measured from 0 downwards, it is
negative. Referring to Figure 16, an angle in
the second quadrant (between 90° and 180 °)
has some of its functions negative:
sin A
cosine, secant
c cos B
a tan B
sec
A =
--
a
c
-a
b
=
neg.
= neg.
-b
cos
pos.
=
pos.
=
neg.
=-a =
neg.
cot A =
c
cosec
b
A =
A
-ba
Summarizing, the sign of the functions in
each quadrant can be seen at a glance from
Figure 17, where in each quadrant are written
the names of functions which are positive;
those not mentioned are negative.
SECOND
QUADRANT
tb
FOURTH
THIRD
QUADRANT
QUADRANT
Figure 16.
TRIGONOMETRIC FUNCTIONS IN THE SECOND, THIRD, AND FOURTH QUADRANTS.
The trigonometric functions in these quadrants are similar to first quadrant values, but
signs of the functions vary as listed in the text and in Figure 17.
www.americanradiohistory.com
the
HANDBOOK
90
A1
270'
180'
Figure 18.
SINE AND COSINE CURVES.
In (A) we have a sine curve
drawn in Cartesian coordinates.
This is the usual representation
of an alternating current wave
without substantial harmonics. In
(B) we have a cosine wave;
note that it is exactly similar
to a sine wave displaced by
90 or n 2 radians.
90"
The sine u ate. When
we have the relation
y= sin x. where x is an
angle measured in radians or degrees, we can
draw a curve of y versus x for all values of
the independent variable, and thus get a good
conception how the sine varies with the magnitude of the angle. This has been done in
Figure 18A. We can learn from this curve the
following facts.
1. The sine varies between +1 and -1
2. It is a periodic curve, repeating itself after
every multiple of 27 or 360°
3. Sin x = sin (180 °
or sin (ir
4. Sin x = -sin (180° + x), or
Graphs of Trigono-
3.
-x)
-sin
srr
360"
--
(R + x)
The cosine :cave. Making a curve for the
function y = cos x, we obtain a curve similar
to that for y = sin x except that it is displaced
by 90° or 7/2 radians with respect to the
Y -axis. This curve (Figure 18B) is also periodic but it does not start with zero. We read
from the curve:
1. The value of the cosine never goes beyond +1 or -1
2. The curve repeats, after every multiple
of 27 radians or 360°
==
=
Cos x
7r
450"
-
--
-
Ado
- - - Now LI
=_
2
= -cos (180°
-x)
-x)
Irr
or
-x)
-x)
r cos (360 °
or cos (tir
The graph of the tangent is illustrated in
Figure 19. This is a discontinuous curve and
illustrates well how the tangent increases from
zero to infinity when the angle increases from
zero to 90 degrees. Then when the angle is
further increased, the tangent starts from
minus infinity going to zero in the second quadrant, and to infinity again in the third quadrant.
1. The tangent can have any value between
4. Cos x
+m
2.
3.
4.
and
-
The curve repeats and the period is err
radians or 180 °, not 2,r radians
Tan x = tan (180° +x) or tan (-rr +x)
Tan x = -tan (180°
or
-tan
-x)
(- -x)
The graph of the cotangent is the inverse of
that of the tangent, see Figure 20. It leads us
to the following conclusions:
1. The cotangent can have any value be-
-
tween
+
2.
It
periodic curve, the period being
3.
4.
radians or 180°
cot (180° +x) or cot (^.r
Cot x
cot (180°
or
Cot x =
is a
-cot
m
and
-
-x)
(r -x)
270°
720'
540°
450
630°
The tangent curve increases from 0 to m with
an angular increase of 90 °. In the next 180'
increases from
-m
to +
180°
630'
450°
270'
90°
0°
360
540°
720'
Figure 20.
COTANGENT CURVES.
Figure 19.
TANGENT CURVES.
it
+x)
en"
360°
180°
C
-
-
_
720'
630°
540
-
z.r
90'
720'
630
"37r
-cos (7
metric Functions
-x)
S
2m
-
o1m
540'
450
360
270"
180"
_
(B)
0°
779
Trigonometric Curves
s-
Cotangent curves are the inverse of the tanin
gent curves. They vary from + m to
each pair of quadrants.
www.americanradiohistory.com
-m
780
THE RADIO
Rodio Mathematics and Calculations
Figure 22.
Vectors may be added as shown in these
sketches. In each case the long vector represents the vector sum of the smaller vectors.
For many engineering applications sufficient
accuracy can be obtained by this method
which avoids long and laborious calculations.
COSINE
Figure 21.
ANOTHER REPRESENTATION OF
TRIGONOMETRIC FUNCTIONS.
If
the radius of a circle is considered as the
unit of measurement, then the lengths of the
various lines shown in this diagram are numerically equal to the functions marked adjacent
to them.
The graphs of the secant and cosecant are
of lesser importance and will not be shown
here. They are the inverse, respectively, of the
cosine and the sine, and therefore they vary
from +1 to infinity and from -1 to infinity.
Perhaps another useful way of visualizing
the values of the functions is by considering
Figure 21. If the radius of the circle is the unit
of measurement then the lengths of the lines
are equal to the functions marked on them.
-
Trigonometric Tables
There are two kinds of
trigonometric tables.
The first type gives the functions of the angles,
the second the logarithms' of the functions.
The first kind is also known as the table of
natural trigonometric functions.
These tables give the functions of all angles
between 0 and 45 °. This is all that is necessary
for the function of an angle between 45° and
90° can always be written as the co- function
of an angle below 45 °. Example: If we had to
find the sine of 48 °, we might write
sin 48° = cos (90° -48 °) = cos 42°
Tables of the logarithms of trigonometric
functions give the common logarithms (log.)
of these functions. Since many of these logarithms have negative characteristics, one should
add -10 to all logarithms in the table which
have a characteristic of 6 or higher. For instance, the log sin 24° = 9.60931 -10. Log
tan 1° = 8.24192 -10 but log cot 1° =
1.75808. When the characteristic shown is less
than 6, it is supposed to be positive and one
should not add -10.
Vectors
A ¡calar quantity has magnitude only; a
vector quantity has both magnitude and direction. When we speak of a speed of 50 miles
per hour, we are using a scalar quantity, but
when we say the wind is Northeast and has a
velocity of 50 miles per hour, we speak of a
vector quantity.
Vectors, representing forces, speeds, displacements, etc., are represented by arrows.
They can be added graphically by well known
methods illustrated in Figure 22. We can make
the parallelogram of forces or we can simply
draw a triangle. The addition of many vectors
can be accomplished graphically as in the same
figure.
In order that we may define vectors algebraically and add, subtract, multiply, or divide
them, we must have a logical notation system
that lends itself to these operations. For this
purpose vectors can be defined by coordinate
systems. Both the Cartesian and the polar coordinates are in use.
Vectors Defined
by Cartesian
Coordinates
Since we have seen how the
sum of two vectors is obtained, it follows from Figure 23, that the vector Z
equals the sum of the two vectors x and y. In
fact, any vector can be resolved into vectors
along the X- and Y-axis. For convenience in
working with these quantities we need to dis-
yS
YA5
o
3
Figure 23.
RESOLUTION OF VECTORS.
i
Any vector such as
may be resolved Into
two vectors, x and y, along the X- and Yaxes. If vectors are to be added, their respective z and y components may be added to
find the x and y components of the resultant
vector.
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HANDBOOK
Vectors
Addition of Vectors
781
An examination of Figwill show that
ure 24
the two vectors
=x,+jY,
Z
may
adding
or
be
added
subtracting
or
their
subtracted
x
or
y
R
by
com-
ponents separately.
tinguish between the X- and Y- component,
and so it has been agreed that the Y- component alone shall be marked with the letter j.
Example (Figure 23)
:
Z
=3 +4j
Note again that the sign of components
along the X -axis is positive when measured
from 0 to the right and negative when measured from O towards the left. Also, the component along the Y -axis is positive when measured from 0 upwards, and negative when
measured from 0 downwards. So the vector,
R, is described as
R
=5 -3j
Vector quantities are usually indicated by
some special typography, especially by using a
point over the letter indicating the vector, as
R.
Absolute Value
of a Vector
The absolute or scalar
value of vectors such as 2
or R in Figure 23 is easily
found by the theorem of Pythagoras, which
states that in any right -angled triangle the
square of the side opposite the right angle is
equal to the sum of the squares of the sides
adjoining the right angle. In Figure 23, OAB
is a right -angled triangle; therefore, the square
of OB (or Z) is equal to the square of OA
(or x) plus the square of AB (or y). Thus the
absolute values of Z and R may be determined
as
follows:
IZI= Vx' +y'
ZI= V3' +4' =5
RI= b +3'=
x:
+
=
Z
j
Y=
+
x,
R
-Z = x, -
The vertical lines indicate that the absolute
or scalar value is meant without regard to sign
or direction.
+
j
+
(y,
Y=)
x:
+
-
(y,
j
y2)
Let us consider the operator j. If we have a
vector a along the X -axis and add a j in front
of it (multiplying by j) the result is that the
direction of the vector is rotated forward 90
degrees. If we do this twice (multiplying by
11) the vector is rotated forward by 180 degrees
and now has the value -a. Therefore multiplying by is equivalent to multiplying by -1.
Then
f
= -1
12
and j
=
V
-1
This is the imaginary number discussed before under algebra. In electrical engineering
the letter j is used rather than i, because i is
already known as the symbol for current.
Multiplying Vectors
When two vectors are
to be multiplied we can
perform the operation just as in algebra, remembering that j2 _ -1.
RZ=
(x,
+j',)
=x,x2
4-
¡XI
=
-
y, Y=
x, x:
+jy2)
(x :
Y:4-jx_yt+j'y,ya
+
j
(x, Ya
+
x:
y,)
Division has to be carried out so as to remove the j -term from the denominator. This
can be done by multiplying both denominator
and numerator by a quantity which will eliminate j from the denominator. Example:
R
x,
+jy,_(x, +jy,)
Z
x:
+
(x:
hY:
+
YiY=
+
x:'
=5.83
x:
For the same reason we can carry out subtraction by subtracting the horizontal components and subtracting the vertical components
x,x:
34
+
can be added, if we add the X- components
and the Y- components separately.
Figure 24.
ADDITION OR SUBTRACTION OF
VECTORS.
Vectors
=
+
j (x:y,
+
(x:
jY:) (x:
-jxa)
- jy:)
- x,y:)
Ya'
A vector can also be defined in polar coordinates by its magnitude and its vectorial angle
with an arbitrary reference axis. In Figure 25
Polar Coordinates
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THE RADIO
Radio Mathematics and Calculations
782
Figure 26.
Vectors can be trans formed from Cartesian
into polar notation as shown in this figure.
X
Figure 25.
IN THIS FIGURE A VECTOR HAS
BEEN REPRESENTED IN POLAR
INSTEAD OF CARTESIAN COORDINATES.
In polar coordinates a vector is defined by
a magnitude and an angle, called the vectorial angle, instead of by two magnitudes
as in Cartesian coordinates.
the vector Z has a magnitude 50 and a vectorial angle of 60 degrees. This will then be
written
i=
5OL6O°
A vector a + jb can be transformed into
polar notation very simply (see Figure 26)
=o +jb= Vo'+ b'Lton'u
In this connection tan -' means the angle of
which the tangent is. Sometimes the notation
arc tan b/a is used. Both have the same meaning.
A polar notation of a vector can be transformed into a Cartesian coordinate notation in
the following manner (Figure 27)
i
=
pLA
=
p cos A
+
jp sin A
A sinusoidally alternating voltage or current is symbolically represented by a rotating
vector, having a magnitude equal to the peak
voltage or current and rotating with an angular
velocity of 2-,rf radians per second or as many
revolutions per second as there are cycles per
which flows due to the alternating voltage is
not necessarily in step with it. The rotating
current vector may be ahead or behind the
voltage vector, having a phase difference with
it. For convenience we draw these vectors as
if they were standing still, so that we can indicate the difference in phase or the phase angle.
In Figure 28 the current lags behind the voltage by the angle e, or we might say that the
voltage leads the current by the angle B.
Vector diagrams show the phase relations
between two or more vectors (voltages and
currents) in a circuit. They may be added and
subtracted as described; one may add a voltage
vector to another voltage vector or a current
vector to a current vector but not a current
vector to a voltage vector (for the same reason
that one cannot add a force to a speed). Figure
28 illustrates the relations in the simple series
circuit of a coil and resistor. We know that the
current passing through coil and resistor must
he the same and in the same phase, so we draw
this current I along the X -axis. We know also
that the voltage drop IR across the resistor is
in phase with the current, so the vector IR representing the voltage drop is also along the
X-axis.
The voltage across the coil is 90 degrees
ahead of the current through it; /X must
therefore be drawn along the Y -axis. E the
applied voltage must be equal to the vectorial
sum of the two voltage drops, IR and IX, and
we have so constructed it in the drawing. Now
expressing the same in algebraic notation, we
have
=IR+jIX
second.
The instantaneous voltage, e, is always equal
to the sine of the vectorial angle of this rotating vector, multiplied by its magnitude.
e
=
E
IZ
Dividing by I
sin 2^rft
The alternating voltage therefore varies with
time as the sine varies with the angle. If we
plot time horizontally and instantaneous voltage vertically we will get a curve like those
in Figure 18.
In alternating current circuits, the current
=
IR
+ jIX
i =R +jX
Due to the fact that a reactance rotates the
voltage vector ahead or behind the current
vector by 90 degrees, we must mark it with a
j in vector notation. Inductive reactance will
have a plus sign because it shifts the voltage
vector forwards; a capacitive reactance is neg-
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HANDBOOK
p
Graphical Representation
COS A
Figure 27.
Vectors can be trans formed from polar into
Cartesian notation as shown in this figure.
ative because the voltage will lag behind the
current. Therefore:
X,.
= +
X
=_j
Graphical Representation
1
2nfC
In Figure 28 the angle 8 is known as the
phase angle between E and I. When calculating power, only the real components count.
The power in the circuit is then
=
P
but IR
OR)
I
=
P=
E
=
tan
8
EIcost)
8
=
Formulas and physical laws are often presented in graphical form; this gives us a
"bird's eye view" of various possible conditions
due to the variations of the quantities involved.
In some cases graphs permit us to solve equations with greater ease than ordinary algebra.
Coordinate Systems
cos
This coi 8 is known as the power factor of
the circuit. In many circuits we strive to keep
the angle e as small as possible, making cos e
as near to unity as possible. In tuned circuits,
we use reactances which should have as low a
power factor as possible. The merit of a coil
or condenser, its Q. is defined by the tangent of
this phase angle:
Q
Figure 28.
VECTOR REPRESENTATION OF A
SIMPLE SERIES CIRCUIT.
The righthand portion of the illustration shows
the vectors representing the voltage drops in
the coil and resistance illustrated at the left.
Note that the voltage drop across the coil Xi.
leads that across the resistance by 90 °.
-fL
j 2
783
X/R
For an efficient coil or condenser, Q should
large as possible; the phase -angle should
then be as close to 90 degrees as possible, making the power factor nearly zero. Q is almost
but not quite the inverse of cos e. Note that in
be as
All of us have used coordinate systems with-
out realizing it. For instance, in modern cities
we have numbered streets and numbered avenues. By this means we can define the location
of any spot in the city if the nearest street
crossings are named. This is nothing but an
application of Cartesian coordinates.
In the Cartesian coordinate system (named
after Descartes), we define the location of any
point in a plane by giving its distance from
each of two perpendicular lines or axes. Figure
30 illustrates this idea. The vertical axis is
called the Y -axis, the horizontal axis is the
X -axis. The intersection of these two axes is
called the origin, O. The location of a point,
P, (Figure 30) is defined by measuring the
respective distances, x and y along the X -axis
and the Y -axis. In this example the distance
along the X -axis is 2 units and along the Yaxis is 3 units. Thus we define the point as
Figure 29
Q
=X /R
and cos
e
=
0= TAN e.
R/Z
When Q is more than 5, the power factor is
than 20%; we can then safely say Q =
/cos e with a maximum error of about 21/2
percent, for in the worst case, when cos 8 =
0.2, Q will equal tan e = 4.89. For higher
values of Q, the error becomes less.
Note that from Figure 29 can be seen the
simple relation:
POWER FACTOR
=COS
e-
less
1
-R+jX,
IZI=
R'+Xt'
Figure 29.
The figure of merit of a coil and its resistance
represented by the ratio of the inductive
reactance to the resistance, which as shown
is
in this diagram is equal to
R'
which equals
tan 9. For large values of 0 (the phase angle)
this is approximately equal to the reciprocal
of the cos O.
-
784
Radio Mathematics and Calculations
Y
2000
7
1800
Ile
SECOND
QUADRANT
FIRST
1600
QUADRANT
5
4
a
1400
1200
1000
2
ROO
600
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100
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5
6
7
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Ill
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5
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Figure 30.
CARTESIAN COORDINATES.
The location of any point can be defined by
its distance from the X and Y axes.
3 or we might say x = 2 and y = 3. The
measurement x is called the abscissa of the
point and the distance y is called its ordinate.
It is arbitrarily agreed that distances measured
from 0 to the right along the X -axis shall be
reckoned positive and to the left negative. Distances measured along the Y -axis are positive
when measured upwards from 0 and negative
when measured downwards from 0. This is
illustrated in Figure 30. The two axes divide
the plane area into four parts called quadrants.
These four quadrants are numbered as shown
P 2,
in the figure.
It follows from the foregoing statements,
that points lying within the first quadrant have
both x and y positive, as is the case with the
point P. A point in the second quadrant has a
negative abscissa, x, and a positive ordinate, y.
This is illustrated by the point Q, which has
the coordinates x = -4 and y = +1. Points
in the third quadrant have both x and y negative. x = -5 and y = -2 illustrates such a
point, R. The point S, in the fourth quadrant
has a negative ordinate, y and a positive abscissa or x.
In practical applications we might draw
only as much of this plane as needed to illus-
trate our equation and therefore, the scales
along the X -axis and Y -axis might not start
with zero and may show only that part of the
scale which interests us.
Representation of
Functions
In the equation:
300,000
200
300
400
600
8
2
IR
THE RADIO
Figure 31.
REPRESENTATION OF A SIMPLE
FUNCTION IN CARTESIAN COORDINATES.
300,000
In this chart of the function fk, =
mr
distances along the X axis represent wavelength in meters, while those along the Y
axis represent frequency in kilocycles. A curve
such as this helps to find values between
those calculated with sufficient accuracy for
most purposes.
f is said to be a function of X. For every value
of f there is a definite value of X. A variable is
said to be a function of another variable when
for every possible value of the latter, or independent variable, there is a definite value of
the first or dependent variable. For instance,
if y = 5x', y is a function of x and x is called
the independent variable. When a = 3b' + 513'
-25b + 6 then a is a function of b.
A function can be illustrated in our coordinate system as follows. Let us take the equation for frequency versus wavelength as an
example. Given different values to the independent variable find the corresponding values
of the dependent variable. Then plot the points
represented by the different sets of two values.
600
800
1000
1200
1400
1600
1800
2000
500
375
300
250
214
187
167
150
Plotting these points in Figure 31 and drawing a smooth curve through them gives us the
curve or graph of the equation. This curve
will help us find values of f for other values
of 7, (those in between the points calculated)
and so a curve of an often -used equation may
serve better than a table which always has
gaps.
When using the coordinate system described
so far and when measuring linearly along both
axes, there are some definite rules regarding
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HANDBOOK
Representation of Functions
785
Figure 32.
Only two points are needed to define functions which result in a straight line as shown
in this diagram representing Ohm's Law.
Figure 33.
TYPICAL GRID - VOLTAGE
PLATE -CURRENT CHARACTERA
the kind of curve we get for any type of
equation. In fact, an expert can draw the curve
with but a very few plotted points since the
equation has told him what kind of curve to
expect.
First, when the equation can be reduced to
the form y = mx + b, where x and y are the
variables, it is known as a linear or first degree
function and the curve becomes a straight line.
(Mathematicians still speak of a "curve" when
it has become a straight line.)
When the equation is of the second degree,
that is, when it contains terms like x' or y'
or Ay. the graph belongs to a group of curves,
called conic sections. These include the circle,
the ellipse, the parabola and the hyperbola. In
the example given above, our equation is of
the form
xy = c,
c being equal to 300,000
which is a second degree equation and in this
case, the graph is a hyperbola.
This type of curve does not lend itself readily for the purpose of calculation except near
the middle, because at the ends a very large
change in represents a small change in f and
vice versa. Before discussing what can be done
about this let us look at some other types of
curves.
Suppose we have a resistance of 2 ohms
and we plot the function represented by Ohm's
Law: E = 21. Measuring E along the X-axis
and amperes along the Y -axis, we plot the
necessary points. Since this is a first degree
equation, of the form y = mx + b (for E
y, m = 2 and / = x and b = 0) it will be a
straight line so we need only two points to
plot it.
-
I
(line passes through origin)
0
5
E
0
10
The line is shown in Figure 32. It is seen to
straight line passing through the origin.
be a
ISTIC CURVE.
The equation represented by such a curve Is
so complicated that we do not use it. Data
for such a curve is obtained experimentally,
and intermediate values can be found with
sufficient accuracy from the curve.
If the resistance were 4 ohms, we should get
the equation E = 4I and this also represents
a line which we can plot in the same figure.
As we see, this line also passes through the
origin but has a different slope. In this illustration the slope defines the resistance and we
could make a protractor which would convert
the angle into ohms. This fact may seem inconsequential now, but use of this is made in the
drawing of loadlines on tube curves.
Figure 33 shows a typical, grid -voltage,
plate -current static characteristic of a triode.
The equation represented by this curve is
rather complicated so that we prefer to deal
with the curve. Note that this curve extends
through the first and second quadrant.
Families of curves. It has been explained
that curves in a plane can be made to illustrate
the relation between to o variables when one
of them varies independently. However, what
are we going to do when there are three variables and two of them vary independently. It
is possible to use three dimensions and three
axes but this is not conveniently done. Instead
of this we may use a family of curves. We
have already illustrated this partly with Ohm's
Law. If we wish to make a chart which will
show the current through any resistance with
any voltage applied across it, we must take the
equation E = IR, having three variables.
We can now draw one line representing a
resistance of 1 ohm, another line representing
2 ohms, another representing 3 ohms, etc., or
as many as we wish and the size of our paper
will allow. The whole set of lines is then
applicable to any case of Ohm's Law falling
within the range of the chart. If any two of
the three quantities are given, the third can be
found.
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Figure 34.
A FAMILY OF CURVES.
fourth variable. But this is not always possible,
for among the four variables there should be
no more than two independent variables. In
our example such a set of lines could represent
power in watts; we have drawn only two of
these but there could of course be as many as
desired. A single point in the plane now indicates the four values of E, I, R, and P which
belong together and the knowledge of any
two of them will give us the other two by
reference to the chart.
Another example of a family of curves is
the dynamic transfer characteristic or plate
family of a tube. Such a chart consists of several curves showing the relation between plate
voltage, plate current, and grid bias of a tube.
Since we have again three variables, we must
show several curves, each curve for a fixed
value of one of the variables. It is customary
to plot plate voltage along the X-axis, plate
current along the Y -axis, and to make different
curves for various values of grid bias. Such a
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Figure 35.
"PLATE"
CURVES FOR A
TYPICAL VACUUM TUBE.
An equation such as Ohm's Law has three
'variables, but can be represented in Cartesian
coordinates by a family of curves such as
shown here. If any two quantities are given,
the third can be found. Any point in the
chart represents o definite value each of E,
1, and R, which will satisfy the equation of
Ohm's Low. Values of R not situated on an R
line can be found by interpolation.
Figure 34 shows such a family of curves to
solve Ohm's Law. Any point in the chart represents a definite value each of E, 1, and R
which will satisfy the equation. The value of
R represented by a point that is not situated
on an R line can be found by interpolation.
It is even possible to draw on the same chart
a second family of curves, representing a
NO
Y
.
we have three variables, plate
In such
voltage, plate current, and grid bias. Each
point on a grid bias line corresponds to the
plate voltage and plate current represented
by its position with respect to the X and Y
axes. Those for other values of grid bias may
be found by interpolation. The loadline shown
in the lower left portion of the chart is explained in the text.
of curves is illustrated in Figure 35. Each
point in the plane is defined by three values,
which belong together, plate voltage, plate
current, and grid voltage.
Now consider the diagram of a resistance coupled amplifier in Figure 36. Starting with
the B- supply voltage, we know that whatever
plate current flows must pass through the
resistor and will conform to Ohm's Law. The
voltage drop across the resistor is subtracted
from the plate supply voltage and the remainder is the actual voltage at the plate, the kind
that is plotted along the X -axis in Figure 35.
We can now plot on the plate family of the
set
+e
Figure 36.
PARTIAL DIAGRAM OF A RESISTANCE COUPLED AMPLIFIER.
The portion of the supply voltage wasted
across the 50,000 -ohm resistor Is represented
in Figure 35 as the loadline.
www.americanradiohistory.com
HANDBOOK
tube the loadline, that is the line showing
which part of the plate supply voltage is across
the resistor and which part across the tube for
any value of plate current. In our example, let
us suppose the plate resistor is 50,000 ohms.
Then, if the plate current were zero, the voltage drop across the resistor would be zero and
the full plate supply voltage is across the tube.
Our first point of the loadline is E = 250,
1 = 0. Next, suppose, the plate current were
1 ma., then the voltage drop across the resistor
would be 50 volts, which would leave for the
tube 200 volts. The second point of the loadline is then E = 200, / = 1. We can continue
like this but it is unnecessary for we shall find
that it is a straight line and two points are
sufficient to determine it.
This loadline shows at a glance what happens when the grid -bias is changed. Although
there are many possible combinations of plate
voltage, plate current, and grid bias, we are
now restricted to points along this line as long
as the 50,000 ohm plate resistor is in use. This
line therefore shows the voltage drop across
the tube as well as the voltage drop across the
load for every value of grid bias. Therefore, if
we know how much the grid bias varies, we
can calculate the amount of variation in the
plate voltage and plate current, the amplification, the power output, and the distortion.
Logarithmic Scales
Sometimes it is convenient to measure along
the axes the logarithm) of our variable quantities. Instead of actually calculating the logarithm, special paper is available with logarithmic scales, that is, the distances measured
along the axes are proportional to the logarithms of the numbers marked on them rather
than to the numbers themselves.
There is semi -logarithmic paper, having
logarithmic scales along one axis only, the
other scale being linear. We also have full
logarithmic paper where both axes carry logarithmic scales. Many curves are greatly simplified and some become straight lines when
plotted on this paper.
As an example let us take the wavelength frequency relation, charted before on straight
cross- section paper.
f
300,000
X
Taking logarithms:
log f
=
log 300,000
-
Logarithmic Scales
-
log
X
If we plot log f along the Y -axis and log X
along the X -axis, the curve becomes a straight
line. Figure 37 illustrates this graph on full
logarithmic paper. The graph may be read
with the same accuracy at any point in con-
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2500
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Figure 37.
A LOGARITHMIC CURVE.
Many functions become greatly simplified and
some become straight lines when plotted to
logarithmic scales such as shown in this
diagram. Here the frequency versus wavelength
curve of Figure 31 has been replotted to conform with logarithmic axes. Note that it is
only necessary to calculate two points in
order to determine the "curve" since this type
of function results in a straight line.
tract to the graph made with linear coordinates.
This last fact is a great advantage of logarithmic scales in general. It should be clear that
if we have a linear scale with 100 small divisions numbered from
to 100, and if we are
able to read to one tenth of a division, the
possible error we can make near 100, way up
the scale, is only 1/10th of a percent. But near
the beginning of the scale, near 1, one tenth of
a division amounts to 10 percent of 1 and we
are making a 10 percent error.
In any logarithmic scale, our possible error
in measurement or reading might be, say 1/32
of an inch which represents a fixed amount of
the log depending on the scale used. The net
result of adding to the logarithm a fixed quantity, as 0.01, is that the anti -logarithm is multiplied by 1.025, or the error is 21/2%. No matter at what part of the scale the 0.01 is added,
the error is always 21/2%.
An example of the advantage due to the use
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1
Radio Mathematics and Calculations
788
10.000
Ï
1.0
0.9
0.8
1000
0.6
0.4
0.3
100
o
IS
10
5
0
5
10
15
KC. OFF RESONANCE
Figure 38.
A RECEIVER RESONANCE CURVE.
This curve represents the output of a receiver versus frequency when plotted to linear
coordinates.
10
9
8
of semi- logarithmic paper is shown in Figures
38 and 39. A resonance curve, when plotted on
linear coordinate paper will look like the curve
in Figure 38. Here we have plotted the output
of a receiver against frequency while the applied voltage is kept constant. It is the kind of
curve a "wobbulator" will show. The curve
does not give enough information in this form
for one might think that a signal 10 kc. off
resonance would not cause any current at all
and is tuned out. However, we frequently have
off resonance signals which are 1000 times as
strong as the desired signal and one cannot
read on the graph of Figure 38 how much any
signal is attenuated if it is reduced more than
about 20 times.
In comparison look at the curve of Figure
39. Here the response (the current) is plotted
in logarithmic proportion, which allows us to
plot clearly how far off resonance a signal has
to be to be reduced 100, 1,000, or even 10,000
times.
Note that this curve is now "upside down ";
it is therefore called a .telectivily curve. The
reason that it appears upside down is that the
method of measurement is different. In a selectivity curve we plot the increase in signal
voltage necessary to cause a standard output
off resonance. It is also possible to plot this increase along the Y-axis in decibels; the curve
then looks the same although linear paper can
4
3
-20
-lo
o
+lo
+20
KC. OFF RESONANCE
Figure 39.
A RECEIVER SELECTIVITY CURVE.
This curve represents the selectivity of a receiver plotted to logarithmic coordinates for
the output, but linear coordinates for frequency. The reason that this curve appears
inverted from that of Figure 38 is explained
in the text.
be used because nuw our unit is logarithmic.
An example of full logarithmic paper being
used for families of curves is shown in the reactance charts of Figures 40 and 41.
An alignment chart con sists of three or more sets
of scales which have been
so laid out that to solve the formula for which
the chart was made, we have but to lay a
straight edge along the two given values on
any two of the scales, to find the third and
unknown value on the third scale. In its simNomograms or
Alignment Charts
www.americanradiohistory.com
,\.-
Figure 40.
REACTANCE -FREQUENCY CHART FOR AUDIO FREQUENCIES
See
text for applications and instructions for use.
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I
!
8
9
o
HANDBOOK
Polar Coordinates
RADIUS
VECTOR
/
/
Figure 42.
THE SIMPLEST FORM OF NOMOGRAM.
plest form, it is somewhat like the lines in Figure 42. If the lines a, b, and c are parallel and
equidistant, we know from ordinary geometry,
that b = 1/2 (a + c). Therefore, if we draw a
scale of the same units on all three lines, starting with zero at the bottom, we know that by
laying a straight -edge across the chart at any
place, it will connect values of a, b, and c,
which satisfy the above equation. When any
two quantities are known, the third can be
found.
If, in the same configuration we used logarithmic scales instead of linear scales, the relation of the quantities would become
log b = 1/2 (log a + log c) orb =
/
PNG-
ORIGIN
o
791
AXIS
X
Figure 43.
THE LOCATION OF A POINT BY
POLAR COORDINATES.
In the polar coordinate system any point is
determined by its distance from the origin
and the angle formed by a line drawn from
it to the origin and the O-X axis.
by the angle A the vectorial angle. We give
these data in the following form
P
=
3
Lo°
Polar coordinates are used in radio chiefly
for the plotting of directional properties of microphones and antennas. A typical example of
such a directional characteristic is shown in
Figure 44. The radiation of the antenna represented here is proportional to the distance of
the characteristic from the origin for every
possible direction.
By using different kinds of scales, different
units, and different spacings between the scales,
charts can be made to solve many kinds of
equations.
If there are more than three variables it is
generally necessary to make a double chart,
that is, to make the result from the first chart
serve as the given quantity of the second one.
Such an example is the chart for the design of
coils illustrated in Figure 45. This nomogram
is used to convert the inductance in microhenries to physical dimensions of the coil and vice
versa. A pin and a straight edge are required.
The method is shown under "R. F. Tank Circuit Calculations" later in this chapter.
Polar Coordinates
Instead of the Cartesian
coordinate system there
is also another system for defining algebraically the location of a point or line in a plane. In
this, the polar coordinate system, a point is determined by its distance from the origin, O,
and by the angle it makes with the axis O -X.
In Figure 43 the point P is defined by the
length of OP, known as the radius vector and
Figure 44.
THE RADIATION CURVE OF AN
ANTENNA.
Polar coordinates are used principally in radio
work for plotting the directional characteristics of an antenna where the radiation is
represented by the distance of the curve from
the origin for every Possible direction.
www.americanradiohistory.com
792
Radio Mathematics and Calculations
Reactance Calculations
In audio frequency calculations, an accuracy
to better than a few per cent is seldom required, and when dealing with calculations involving inductance, capacitance, resonant frequency, etc., it is much simpler to make use of
reactance -frequency charts such as those in
figures 40 and 41 rather than to wrestle with a
combination of unwieldy formulas. From these
charts it is possible to determine the reactance of
a condenser or coil if the capacitance or inductance is known, and vice versa. It follows from
this that resonance calculations can be made
directly from the chart, because resonance
simply means that the inductive and capacitive
reactances are equal. The capacity required to
resonate with a given inductance, or the inductance required to resonate with a given capacity, can be taken directly from the chart.
While the chart may look somewhat formidable to one not familiar with charts of this
type, its application is really quite simple, and
can be learned in a short while. The following
example should clarify its interpretation.
For instance, following the lines to their intersection, we see that 0.1 hy. and 0.1 pfd. intersect at approximately 1,500 cycles and 1,000
ohms. Thus, the reactance of either the coil or
condenser taken alone is about 1000 ohms, and
the resonant frequency about 1,500 cycles.
To find the reactance of 0.1 hy. at, say,
10,000 cycles, simply follow the inductance
line diagonally up towards the upper left till it
intersects the horizontal 10,000 kc. line. Following vertically downward from the point of
intersection, we see that the reactance at this
frequency is about 6000 ohms.
To facilitate use of the chart and to avoid
errors, simply keep the following in mind: The
vertical lines indicate reactance in ohms, the
horizontal lines always indicate the frequency,
the diagonal lines sloping to the lower right
represent inductance, and the diagonal lines
sloping toward the lower left indicate capacitance. Also remember that the scale is logarithmic. For instance, the next horizontal line
above 1000 cycles is 2000 cycles. Note that
there are 9, not 10, divisions between the heavy
lines. This also should be kept in mind when
interpolating between lines when best possible
accuracy is desired; halfway between the line
representing 200 cycles and the line representing 300 cycles is not 250 cycles, but approximately 230 cycles. The 250 cycle point is approximately 0.7 of the way between the 200
cycle line and the 300 cycle line, rather than
halfway between.
Use of the chart need not be limited by the
physical boundaries of the chart. For instance,
the 10 -µpfd. line can be extended to find where
it intersects the 100 -hy. line, the resonant frequency being determined by projecting the intersection horizontally back on to the chart.
To determine the reactance, the logarithmic
ohms scale must be extended.
When winding coils for use in
radio receivers and transmit ters, it is desirable to be able to
determine in advance the full
coil specifications for a given frequency. Likewise, it often is desired to determine how much
capacity is required to resonate a given coil so
that a suitable condenser can be used.
Fortunately, extreme accuracy is not required, except where fixed capacitors are used
across the tank coil with no provision for trimming the tank to resonance. Thus, even though
it may be necessary to estimate the stray circuit capacity present in shunt with the tank
capacity, and to take for granted the likelihood
of a small error when using a chart instead of
the formula upon which the chart was based.
the results will be sufficiently accurate in most
cases, and in any case give a reasonably close
point from which to start "pruning."
The inductance required to resonate with a
certain capacitance is given in the chart in
figure 41. By means of the r.f. chart , the
inductance of the coil can be determined,
or the capacitance determined if the inductance is known. When making calculations, be
Tank
Circuit
Colculotions
R. F.
sure to allow for stray circuit capacity, such as
tube interelectrode capacity, wiring, sockets,
etc. This will normally run from 5 to 25 micro microfarads, depending upon the components
and circuit.
To convert the inductance in microhenries
to physical dimensions of the coil, or vice
versa, the nomograph chart in figure 45 is
used. A pin and a straightedge are required.
The inductance of a coil is found as follows:
The straightedge is placed from the correct
point on the turns column to the correct point
on the diameter -to- length ratio column, the
latter simply being the diameter divided by the
length. Place the pin at the point on the plot
axis column where the straightedge crosses it.
From this point lay the straightedge to the correct point on the diameter column. The point
where the straightedge intersects the inductance column will give the inductance of the
coil.
From the chart, we see that a 30 turn coil
having a diameter -to- length ratio of 0.7 and a
diameter of 1 inch has an inductance of approximately 12 microhenries. Likewise any one
of the four factors may be determined if the
other three are known. For instance, to determine the number of turns when the desired in-
www.americanradiohistory.com
COIL CALCULATOR NOMOGRAPH
Figure 45.
For single loyer solenoid coils, any wire size. See
N
PLOT
OF
AXIS
TURNS
-
-
text for instructions.
RATIO DIAMETER
LENGTH
865-
400
- 20000
300
4
-10000
-
200
150
-100
-
INDUCTANCE IN
MICROHENRIES
90
80
70
80
50
--
--
-4
4000
3000
2-
-3
1000
800
600
400
300
-2
200
--
-S
3-
- 2000
-40
8000
6000
-
DIAMETER
INCHES
100
80
60
40
30
20
.8-
--
- --
-1.5
.8-
1Ó
8
-
r- 6
20
--4
3
-15
-
10
2
- .6
-.75
-.4
-.3
-.2
-5
-- o
--4
3
www.americanradiohistory.com
794
Radio Mathematics and Calculations
D/L ratio, and the diameter are
known, simply work backwards from the example given. In all cases, remember that the
straightedge reads either turns and D/L ratio,
or it reads inductance and diameter. It can
read no other combination.
The actual wire size has negligible effect
upon the calculations for commonly used wire
sizes (no. 10 to no. 30). The number of turns
of insulated wire that can be wound per inch
(solid) will be found in a copper wire table.
ductance, the
Significant Figures
In most radio calculations, numbers represent quantities which were obtained by measurement. Since no measurement gives absolute
accuracy, such quantities are only approximate
and their value is given only to a few significant figures. In calculations, these limitations
must be kept in mind and one should not finish for instance with a result expressed in more
significant figures than the given quantities at
the beginning. This would imply a greater accuracy than actually was obtained and is therefore misleading, if not ridiculous.
An example may make this clear. Many ammeters and voltmeters do not give results to
closer than 1/4 ampere or 1/4 volt. Thus if we
have 21/4 amperes flowing in a d.c. circuit at
63/4 volts, we can obtain a theoretical answer
by multiplying 2.25 by 6.75 to get 15.1875
watts. But it is misleading to express the answer down to a ten -thousandth of a watt when
the original measurements were only good to
1/4 ampere or volt. The answer should be expressed as 15 watts, not even 15.0 watts. If
we assume a possible error of 1/8 volt or ampere (that is, that our original data are only
correct to the nearest 1/4 volt or ampere) the
true power lies between 14.078 (product of
21/8 and 65/8) and 16.328 (product of 23/8 and
67/8). Therefore, any third significant figure
would be misleading as implying an accuracy
which we do not have.
Conversely, there is also no point to calculating the value of a part down to 5 or 6 significant figures when the actual part to be used
cannot be measured to better than 1 part in
one hundred. For instance, if we are going to
use 1% resistors in some circuit, such as an
ohmmeter, there is no need to calculate the
value of such a resistor to 5 places, such as
1262.5 ohrr>i. Obviously, 1% of this quantity
is over 12 ohms and the value should simply
be written as 1260 ohms.
There is a definite technique in handling
these approximate figures. When giving values
obtained by measurement, no more figures are
given than the accuracy of the measurement
permits. Thus, if the measurement is good to
two places, we would write, for instance, 6.9
which would mean that the true value is
somewhere between 6.85 and 6.95. If the measurement is known to three significant figures,
we might write 6.90 which means that the true
value is somewhere between 6.895 and 6.905.
In dealing with approximate quantities, the
added cipher at the right of the decimal point
has a meaning.
There is unfortunately no standardized system of writing approximate figures with many
ciphers to the left of the decimal point. 69000
does not necessarily mean that the quantity is
known to 5 significant figures. Some indicate
the accuracy by writing 69 x 10' or 690 x 10'
etc., but this system is not universally employed. The reader can use his own system, but
whatever notation is used, the number of significant figures should be kept in mind.
Working with approximate figures, one may
obtain an idea of the influence of the doubtful
figures by marking all of them, and products
or sums derived from them. In the following
example, the doubtful figures have been underlined.
603
34.6
0.120
637.720
answer:
638
Multiplication
654
0.341
654
0.342
19612
26116
1308
2616
1962
223.668
1108
answer: 224
224
It is recommended that the system at the
right be used and that the figures to the right
of the vertical line be omitted or guessed so as
to save labor. Here the partial products are
written in the reverse order, the most important ones first.
In division, labor can be saved when after
each digit of the quotient is obtained, one figure of the divisor be dropped. Example:
www.americanradiohistory.com
1.28
527
1
673
527
53
146
106
5
40
40
INDEX
795
www.americanradiohistory.com
A
414
92
694
734
Absorption, Ionospheric
Acceptor, def. of
A.C. -D.C. power supply
A -C V -T voltmeter
Adaptor, F -M
Admittance, def. of
"A" -frame antenna mast
Air capacitor
Air dielectric
Air -gap, capacitor
B.f.o.
Filter
Alignment
"
- - ----
Receiver
I
-F
All- driven antenna
Alpha, def. of
Alternating current
"
444
355
32
262
Amplitude
Effective value
Def. of
Generation
-
Amateur frequency bands
Amateur Radio
Ammeter
Ampere, def. of
"
-
D.C.
"
"
"
47
45, 46
45
61
59
52
42
12
11
731
22, 23
36, 62
73, 74
678
225, 212
169
168
27, 256
127
108, 118
108, 137
129, 249
124, 249
118, 123
117
613
296
126
110, 111
129
1
99,
AB1
99, 108, 123,
108,
113,
inductive
distortion products
"
93
45
Average
41, 45
42
--
coupling
"
"
"
"
"
turns
Class AB
Class B
Class C
"
500
Amplification factor
Amplifier
audio, 15 -watt
" Audio
Cascode
cascode grounded -grid
constant current curves
--
"
"
licenses
Cathode driven
Cathode
Cathode follower
A2
Class A
Response
R.F., Class AB1
R.F., Class B
234
233, 234, 180
Impedance
"J" operator
Phase angle
Ohm's law
R.M.S.
Pulsating
Transformer
Transient
Voltage divider
Alternator
Push -pull
"
"
321
52
R -F
"
Doherty
Driver
Equivalent circuit
feedback
121
Grid circuit
162
Grounded cathode, grounded screen
162, 212, 256, 612
Grounded -grid
612
Grounded -grid, use of
617
350 w., grounded -grid
627
813 grounded -grid
634
KW -2 grounded -grid
146
Hi -fi, 25 -watt
138, 112
High mu (A)
Hi -fi
171, 208
I -F
Horizontal
649
Kilowatt
649
Kilowatt 4 -400A
284
Linear
654
Linear, 2kw. P.E.P.
166
Load line
118
Loftin -White
250, 615
Neutralization
152
Noise factor
199
Operational
120
Pentode
609
Pi- network
159
Plate efficiency
18, 602
Power
"
"
"
-F
R
-F
112
166
168
611, 612
Grounded screen
RC
109
4CX -1000A
211
Semi -conductor
104
200
201
296
Summation
Summing
Terman- Woodyard
Tetrode
Transistor
606
104
"Tri- band" 4CX300A
Ultra- linear
Vertical
Video
V.H.F.
622
s
142
138
112, 113, 128
"strip -line"
"
Voltage
" Williamson
" 3 -10002
" 4 -400A all -band
" 4CX1000A, 2 kw.
Amplitude
A.C.
Amplitude, Distortion
Modulation..109,
Analog computers
problems
"And -or" circuitry
Angle of radiation
V.h.f.
407,
Anode, def. of
dissipation
Antenna
Adjustment
" All- driven
" Bandwidth
Beam
design chart
2- element
Bidirectional
Bi- square
Bobtail
Broadside
-
-
---
- - -
Bruce
Center -fed
Colinear
Combinations of
Construction
Corner reflector
Couplers
mobile
-
Coupling systems
Cubical quad
Curtain
Delta match
Dipole array
Directivity
-
broad -band
Discone
Doublet
Dummy
Efficiency
Element spacing
-
- -
1
--
R.F.,
R
121, 604
End -effect
End -fed
end -fire
Feed systems
Franklin array
Fuchs
Gain, beam
140
661
643
654
45
280, 669
195, 200
204
456, 474
67, 72
505
500
401, 409
494, 491
501, 466
468
464
466
434
462
471
444, 502
481
450, 516, 520
447
467
464
497
461, 431
400, 406
437, 477
423
736
405
464
402
422, 433, 469
424, 494
462
422
491
499
405
427
422
455
404
473
Lazy -H
464
length
Length -to- diameter ratio_.401, 402
Long wire
457
Gamma
t.
595
110
match
Ground loss
Ground plane
Hertz
High frequency type
Impedance
Intercept, v.h.f.
-
796
www.americanradiohistory.com
"
"
"
"
"
"
"
"
"
"
"
"
to
vt
It
1.
Marconi
Matching systems
Measurements
Multee
Multi -band
Mutual coupling
Parasitic beam
404, 428
438
740
436
432, 510
475
490
400, 456
462
400
475
Patterns
Phasing
Polarization
Polarization, v.h.f.
Power gain
0
401
-
405
404, 400
460
490, 498
Reactance
Resonance
Rhombic
Rotary
Rotary match
-
Rotator
Single wire feed
Six- shooter
Sleeve
Space conserving
Stacked
Sterba
Stub adjustment
T -match
3- element beam
508
437
468
476
-
stub match
430
464, 494
465
441, 499
497
484
tt
Triplex
471
to
Tuner
452
476
434
458
426
to
ft
el
0.
O.
OS
tt
to
Turnstile
Two -band Marconi
V-type
Vertical
V.h.f.
8 element, ground plane....484,
479,
V.h.f.
helical
horn
V.h.f. nondirectional
V.h.f. rhombic
487,
V.h.f.
Yagi
Screen array
W8JK
X -array
--
-
-
Yoke match
t.
"Zepplin"
Antennascope
Antennascope, SWR meter
Anti- resonance
Aquadag
Area, capacitor plate
Arrays, antenna
"Assumed" voltage
atomic number
Atom, def. of
-
-
Attenuation, Bass /Treble
Amplifier
Audio
" 15 watt
" Distortion
Equalizer
Feedback loop
Filter, SSB
Frequency, def. of
Hum
limiters
Phase Inverter
-
SSB
Phasing network
Rectification
Wiring technique
Autodyne detector
Automatic load control, SSB
Automatic modulation control
Automatic volume control
Autotransformer
Avalanche voltage
477
482
478
482
484
469
465
494
423, 463
748
747
54
85
33
461, 408
52
21
139
225
678
135
139
141
329
42
142
155, 185, 226
115
334
379
142
206
341
670
223
63, 386
697
-
B
Balanced modulator
SSB
327, 334, 339
350
Balanced modulator, deflection
746
Balanced SWR bridge
744
Balanced transmission line measurements
12
Bands, amateur
215
Bandspread tuning
401, 409
Bandwidth, antenna
Bandwidth, modulation
Base electrode
Bass suppression, audio
Battery bias
Beam
deflection modulator
" Design chart
Dimensions
" Element spacing
Front-back ratio
H -F
5- element
2- element
-
-
-
-
Power tube
-
492
206
222
36
-H curve
Bias
"
267
266
Battery
Cathode
Def. of
cut -off
Grid
Grid leak
R
-
-F
73
108, 265
112, 266
604
266
305, 307
675
amplifier
Safety
Shift modulator
Supply, modulator
Supply, regulated
Transistor
Bidirectional antenna
Billboard antenna
Binary notation
Bishop noise limiter
Bi- square antenna
Bistable multivibrator
Bit
71 2
96
501
473
195, 196
226
466
102
196
376
27, 709, 391
394
397
102, 189, 190
468
70
139
203
737
738
716
689, 690
738
-
Blanketing interference
Bleeder resistor
safety
Blocked grid keying
Blocking diodes
Blocking oscillator
Bob -tail antenna
Bombardment, cathode
Boost, Bass /treble
Break voltage
Bridge, impedance
" Measurements
" Power supply
" Rectifier
" Slide -wire
"
92
304
267
350
494, 492
493
492
492, 490
78
491
Radiation resistance
" Three element HF
220 Mc.
Beat note
Beat oscillator
B
281
454, 746
SWR
Bridge -T oscillator
Bridge -type vacuum tube voltmeter
Bridge, Wheatstone
Broad band dipole
Broadcast interference
Broadside antennas
arrays
Bruce antenna
Butterfly circuit
-
-
C
Calorimeter
Capacitance
Calculation
" Interelectrode
" Neutralization
" Tank
Stray
797
www.americanradiohistory.com
-
definition
192
131
738
431
375
464, 408
466
231
736
32, 30
76, 106
107
259, 216
268
46
Capacitive coupling
Capacitive reactance
A.C. circuit
Capacitor
" Air
Characteristics
" Breakdown
Charge
Color code
Electrolytic
Equalizing
Filter
--
Clipping, negative peak
Clipping, phase shift
Clipping, speech
34
355, 354
Closed loop feedback
262
Coaxial reflectometer
Coaxial transmission line
Coaxial tuned circuit
31
527
34, 708
Cade
Code,
34
707, 690
30
Fixed
Input filter
Leakage
Low inductance
Parallel
714, 690
Coercive force
Coil, core loss
Coil, Q
34
611
33
55
33
32
39
Q
Series
Colinear antenna
Collector electrode
Collins feed system
Collins filter
Color code, component
Color code, standard
Colpitts oscillator
Complementary symmetry
Complex quantity
Component color code
Component nomenclature graph
Compound, def. of
Temperature coefficient
Time constant
Vacuum
variable vacuum
360
Voltage rating
34
Distortion
Carrier
309
282
Carrier, Shift
Carrier, S.S.B.
323, 327, 328, 330
V.H.F.
111, 212
Cascade amplifier
Bias
Cathode
108, 266
115, 116
" Coupling
coupled inverter
Definition
" Current
79, 67
" Driven amplifier
612, 256
Follower
272
Follower amplifier
127, 164
Follower driver
680, 683
Follower modulator
288
Keying
393
Modulation
295
Ray oscilloscope
tube
170, 84
Tank, grounded -grid
165
Cavity, resonant
230
Cell, Weston
24
Center -fed antenna
434
Ceramic dielectric
32
Charge
22, 23, 30
-
- --
Compression, volume
Compressor, A.L.C.
Computers, electronic
Conductance, def. of
Conduction
Conductor
Conductivity
Constant amplitude recording
Constant current curve
Constant -K filter
Constant velocity recording
Constants, vacuum tube
Control circuit, mobile
Control circuit, transmitter
Conversion conductance
Conversion frequency, SSB
Converter stage
Converter, transistor
-
-
Chassis layout
Choke, filter
Choke input filter
Choke, R -F
Circuit constants, measurement of
Circuits
Coupled
" D.C.
Equivalent
" Input loading
" Limiting
Magnetic
Parallel
Parasitic
--
Q
1.
Resonant
- Series
Circulating current
Clamping circuit
Clapp oscillator
Class
Class
Class
Class
Class
Class
Class
Class
Class
Class
amplifier, def. of
A2 amplifier
AB amplifier, def. of
ABI amplifiers
B amplifiers
B amplifier, grounded -grid
B modulator
C
R -F amplifier
amplifiers
C
bias
B,
Clipper -Amplifier design
Clipper filter, speech
Clipping circuits
726
709
713, 715
270, 357
737
Core loss
Corner reflector antenna
Corona static
Coulomb
Counter e.m.f
Counting circuit
Capacitive
Coupling
" Cathode
Choke
Critical
Direct
Effect of
Impedance
Inductive
Interstage
r -f
Link
153
21, 51
152
185
-
35
25
361
55
--
Tank ....53, 24, 55, 57
56
187, 188
A
Magnetic
239
108
118
108
108
108, 123, 125, 159
613
126, 292
249
108, 124, 125, 249
267
678
302
practice set
Coefficient of coupling
Coefficient, temperature
Parasitic
R -F
feedback
Systems, antenna
Transformer
670
299
298, 670
192
742
419
230
14 -20
20
38
32
37
55
54, 356
462
92
443
220
528
527, 528
239
97
49
528
526
21
670
341
194
52
67, 90
23
137
74, 157, 168
64
136
107
521
387
80
336
209
100
55
481
525
22, 31
37
190
268
115
114
57, 216
115
56
113, 269
268
270, 603
38
360
273
447
113
Unity
Critical coupling
Critical inductance
Cross modulation
Crossover point
269
216
686
377
136
C.R.P.L. Predictions
414
244
218, 332
244
Crystal
"
"
185
--
Current
Filter
Lattice filter
Harmonic
798
www.americanradiohistory.com
-
"
Oscillator
Quartz
" Pickup
Cubical quad antenna
Alternating
Current
" Amplification (alpha)
" Cathode
Circulating
Closed path
Def. of
" Effective
Effective (a.c.)
-
"
"
"
"
93
79
-
56, 28
106
42, 44
693
731, 733
125
amplifier
-
"
Rating, Power supply
Saturation
Skin effect
Curtain antenna
"
414
D
D. C.
clamping circuit
D.C. power supplies
D.C. restorer
circuit
Defector
autodyne
Deflection, plate
Degeneration, transistor
Degree, electrical
Delta match
antenna
Demodulator
D. C.
-- - -
187
684
188
206
crystal
-
--
96
43
system....497, 426, 438
205
-
-
1
329, 349
256
258
423, 425
93
126
680, 683
411
726
95
522
240
E
171
208
Detection, slope
Detection, synschronous, DSB
349
222
Detector
Diode
208
" Fremodyne
222
Grid leak
Impedance
Plate
Product
235, 348
Ratio
319
207
Super -regenerative
Deviation, FM
Measurement
310, 316
Dielectric, ceramic
Constant (K)
30, 31, 32
Differential keying
396
198
Differentiation, electronic
Differentiator (RC)
59
Digital circuits
195
computers
Digital package
204
Diode
A.v.c.
223
" Blocking
397
232
Crystal
Def. of
71
Detector
222
1.
204
Gate
Limiter
85
Mixer
210
Modulator
328
90
Semi -conductor
103
Storage time
Voltmeter
734
105
Zener
Dipoles
399, 432, 461, 494
Direct current circuits
21
455
Directive antennas, H -F
Directivity, antenna
400, 406
30
Discharge of capacitor
Discone antenna
437, 477
421
Discontinuities, transmission line
318
Discriminator, F -M
-
98
190
26
296
334
213
-
731
117
amplifier
109, 135
613
340
135
-
41
d'Arsonval meter
309
119
136
305
Transient
Transistor
Divider, frequency
Divider, voltage
Doherty amplifier
Dome audio phasing network
Double conversion circuit
Double sideband, DSB
Doubler, frequency
Doubler, push -pull
multi -wire
Doublet, antenna
Drift transistor
Driver, amplifier
Driver, cathode follower
Duct propagation
Dummy antenna
dummy loads
Dynamic resistance
Dynamotor, PE -103
Dynatron oscillator
93
73
285
64
Cycle
Cycle, sunspot
Frequency
Products
Products, SSB
685
72, 55
464
Cutoff (alpha)
Cutoff bias
Cutoff, extended
Cutoff frequency
72
165
109
135
-
22
733, 45
Electrode
Induced
Instantaneous
Inverse
Measurement
-
-
41
-
Peak
Dissipation, anode
Dissipation, grid
Distortion
Amplitude
" Audio
" Carrier
" Harmonic
Intermodulation
Modulation
Phase
Nonlinear
242, 243
137
467
38
Eddy current
Efficiency, antenna
Electric filters
Electrical energy
Electrical potential
405
63
29
22
Electromagnetic deflection
Electrolytic capacitor
Electrolytic conductor
Electromagnetism
Electromotive force (e.m.f.)
Electron coupled oscillator
drift
orbit
Electron, def. of
Electronic computers
Electronic conduction
integration
Electronic differentiation
Electronic Keyer, "910"
Electronic multiplication
Electrostatics
Electrostatic deflection
energy
metallic, non -metallic
Element, def. of
Element, reactive, non- reactive
Emission equation
photoelectric
Emission
" Secondary
" Spurious
" Thermionic
Emitter electrode
- - -
-
Enclosures
End effect, antenna
End -fed antenna
End -fire antennas
-
--
arrays
Electrostatic
Electrical
Energy
" Potential
" Storage, capacitor
" Transistor
ENIAC
Envelope, carrier
Equalizer, audio
Equal- tempered scale
Equivalent circuit
799
www.americanradiohistory.com
86
34, 708
22, 67
35
22, 37
239
67, 21
194
67
198
597
198
30
83, 30
21, 90
21
70
67
71, 77
379
67
92
724, 725
402
422, 433
469, 408
29, 30
30
31
93
195
280
139
135
51, 110, 111
Equivalent circuit, transistor
Equivalent noise resistance
Error signal
Excitation, grid
Exciters, low power
Exciters, SSB
Extended -Zepp antenna
Pulse repetition
95
153
193
250
576
342, 345
463
Shift keying
Sound
Spectrum
Spotter
Standard WWV
Sweep
Frequency Modulation
F
Factor of merit
Fading
Farad, def. of
Feedback amplifier
54
414
Audio
141
"
"
Circuits
Control
Error cancellation
192
192
193
199
"
"
"
"
31
Miller
Transistor
Feed systems, antenna
Feedthrough power
Field, magnetic
Filament, def. of
Filament reactivation
Filter
Capacitor
" Capacitor input
" Carrier, SSB
" Choke
Choke input
Circuits, mobile
Crystal
Crystal lattice
Generator, SSB
High Pass
Inductor input
Insertion loss
101
-
713, 715
514
SSB
218
332
330
64, 368
687
65
64, 372
M- derived
64
Mechanical
220, 332
Noise
225
Passband, SSB
332
O- multiplier
234
Resistance- capacitance-Resonance -.687, 688
Ripple factor
688
Sections
Series
Shunt
64
TVI, receiver
TVI -type
65, 372
Wave
63
Filter -type exciter, SSB
345
Fixed bias
108
Flat -top beam antenna
469
Fletcher- Munson curve
140
Floating Paraphase inverter
116
Flux, def. of
density
35, 36
Flywheel effect
57, 257
Folded dipole
440, 425, 494
Forcing function
200
Foster -Seely discriminator
319
Franklin antenna
462
Franklin oscillator
241
Free electrons
21
Free -running multivibrator
189
Fremodyne detector
208
Frequency
41, 58
" Conversion, SSB
336
" Cutoff
64
" Distortion
109
Divider
190
-- -
-
Multiplier
207
413
739
256
321
index
310
318
320
314
311
321
317
492
422
antenna
Full -wave limiter
Full -wave rectifier
612
35
67
68
707, 690
714, 690
330
709
low pass
Interruption
Maximum useable
Measurements
Fuchs
424, 494
-
- -
740
739
172
308
Reception
Front -back ratio, beam
272, 607
R -F
"
Adapter
41
Deviation
ratio
Discriminator
Limiter
Linearity
Narrow band
Pre -emphasis
129
"
190
136
176
53
322
134
Range, hi- fidelity
Ratio, Lissajous
Resonant
Function, Forcing
Function generator
228
-
non -linear
-
689
200
ramp --202, 203
G
-- - --
G, conductance, def. of
Gain, antenna
beam
Gain, power
resistance
voltage
Gamma match, antenna
system
Gas tube
generator
Gate, diode
limiter
Gauss, def. of
Generator, function
Generator noise
52
491, 457
94
499, 440
87, 172
204, 185
36
202
524, 750
Generator, sawtooth
172
Generator, time base
171
Gilbert, def. of
36
Grid bias
108, 265
Grid, def. of
72
Grid dissipation
165
Grid excitation
250
Grid leak bias
109, 112, 266
Grid leak detector
222
Grid limiter, limiting
187
Grid modulation
250, 286
Grid neutralization
251
Grid-screen
factor
78
Ground bus
142
Ground currents
358
Ground loss, antenna
405
Ground plane antenna
VHF
427, 477
Ground resistance
406
Ground, R -F
358
Ground termination
429
Ground, transmitter
390
Ground, wave
410
Grounded -cathode amplifier
162
Grounded -grid amplifier
162, 212, 256, 612
Grounded -grid cascade amplifier
169
Grounded -grid, cathode tank
165
Grounded -grid r.f. amplifier
612
Guy wires, antenna
445
-
H
Hairpin coupling
Half -wave rectifier
Harmonic
B.f.o.
" Crystal
" Def. of
-
800
www.americanradiohistory.com
230
689
223
244
58
"
"
"
"
Distortion
119
Initial condition voltage
203
Music
135
Injection voltage
Input loading
Input resistance
Insertion loss, filter
211
Oscillator
Radiation
Harmoniker TVI filter
Hartley oscillator
Hash rectifier
245
260, 369
375
238
694
353
99
104, 707
70
292
479
37
422
206
455
134 -138
382
368
Heat cycle, resistor
Heat sink
Heat sink, transistor
Heater cathode
Heising modulation
Helical antenna
Henry, def. of
Hertz antenna
Heterodyne
H -F antennas
High fidelity
High fidelity, Interference
High -pass TVI filter
Holes in semi -conductor
91
Horizontal directivity
Horn antenna, VHF
Hot cathode phase inverter
Hum, audio
Hysteresis loop
loss
406
482
116
-
142
36, 38
I
-f alignment
180, 233
" Amplifier
208
" Noise limiter
225
" Pass -bond
217
Rejection notch
220
" Shape factor
217
" Tuned circuit
216
Ignition noise
523
Images -image interference- ratio....209, 211, 381
Impedance
46, 47
" Antenna
404
" Bridge
737
" Complex
51
" Coupling
56, 113
" Match, audio
127
" Reflected
56, 63
" Resonant
54
" Screen circuit
287
I
"
"
260
Transformation
63
97
" Transistor
" Transmission line
417
" Triangle
48
742
Incident voltage, transmission line
748
Indicator, Antennascope
Indicator, Selsyn
510
Indicator
standing wave
twin- lamp..741, 745
Induced current
42
37
Inductance
" Capacitor
355
" Cathode lead
153
" Critical
686
" Lead
Magnetic
Mutual
81, 37
" Parallel
Series
38
" Resistor
353
" Screen lead
255
Induction, def. of
42
Inductive reactance
46
Inductive coupling
269
Inductive tuning
609
Inductor, iron core
38
Inductor, time constant
40
Infinite impedance detector
222
"
-
-
--
Instability, R -F amplifier
Instability, static
Insulation, VHF
Insulator, def. of
Integration amplifier
Integration, electronic
Integrator, Miller
Integrator (RC)
-
607
117
475
22, 23
200
198
199
- -
Interelectrode capacitance
Interference
Broadcast
" Harmonic
" Hi -fi
image
TV
Interlock, power
Intermodulation distortion
Intermodulation test
Internal resistance
International Morse Code
Interruption frequency
Interstage coupling
Intrinsic semi -conductor
Inverse current
Voltage
Inverter, phase
voltage divider
Ion, def. of
Positive
Ionosphere, absorption
Ionosphere, Sporadic -E layer
Ionospheric cycle
Ionospheric fading
Ionospheric layers
propagation
Ionospheric reflection
IR drop
-
--
-
59
76
375
369
381, 382, 367
391
136
145
25
15
207
268
91
115,
1
693
16, 117
22, 87
13, 414
414
414
414
412, 413
415
24
Iron vane meter
733
Isotropic radiator
406
J
Jitter
Johnson
'J"
0 -feed
188
442
system
Operator
47
Joule, def. of
Junction, transistor
30, 37
93
K
420
Surge
Tank circuit
152
107, 214, 274
65
dielectric constant
32
Key clicks
Keyer, "9T0" electronic
Keying
blocked grid
Cathode circuit
392
K,
-
"
597
394
393
396
322
399
395
42
Differential
Frequency shift
Screen grid
Transmitter
Kilocycle
Kinescope tube
Kirchhoff's Laws
Kylstron
reflex
84
-
28
81, 82
L
Lamb noise limiter
Layout, chassis
Lazy -H antenna
L/C ratio
Lead inductance
Leakage, capacitor
Leakage reactance
Left -hand rule
Length, antenna
L- section filter
801
www.americanradiohistory.com
226
726
464
56
81
34
63
35
401
64
audio
diode
226
Mechanical filter, use of
Megacycle
185
Megohm
-M
full -wave
noise
series diode
320
228
225
Memory circuit
Mercury vapor rectifier
Meteor bursts
Meter, d'Arsonval
Meter, iron vane
Meter, multi -range
Meter, rectifier
Meter shunt
Meter, thermocouple
Mho, def. of
Mica dielectric
Microfarad
Microhenry
Micro -ohm
Middle C
Miller effect
Miller feedback
oscillator
Miller integrator
Milliammeter
Millihenry, def. of
Mixer
circuits
" Diode
" Noise
" Products, SSB
spurious
12
Licenses, amateur
Limiter,
Limiter,
Limiter,
Limiter,
Limiter,
Limiter,
Limiter,
Limiting
Limiting
F
185
228
185, 186, 202
204
INS
circuit
diode
Line filter
Line regulation
Line, Slotted
Linear amplifier
"
Class
"
Kilowatt
B
"
"
Tuning
2KW, P.E.P.
Linear matching transformer
Linearity, distortion products
Linearity tracer
antenna
Link coupling
Lissajous figures
Litz, wire
L- network design
Load line
Load line, r.f. amplifier
Load line, transistor
Load resistance amplifier
Load, r.f. dummy
loaded -O
Loftin -White amplifier
Long -wire antenna
Loop feedback
Loran, band
Loudness control, audio
response
Loudspeaker
def. of
Low- frequency parasitics
Low -pass TVI filter
-
-
-
227
384
740
284
129
649
285
654
442
613
185
270, 448
176
55
263
74
166
99
119
736
57
118
457
192
12
140
139, 140
362
372
M
-- -
Magic eye tube
Air gap
Magnetic
field
flux
" Circuit
" Eddy current
" Hysteresis loss
" Induction
" Left hand rule
reluctance
" Permeability
residual
Magnetism
Magnetomotive force (M. M. F.)
- -
Magnetron
Magnitude, scalar
Majority carrier
Marconi antenna
Mast, A -frame
Matching stub, antenna
Matching systems antenna
Matching transformer
Mathematics, radio
Maximum usable frequency
Maxwell, def. of
-
M- derived filter
Antenna
Measurements
Bridge
Circuit constants
Voltage
Current
Frequency
Parallel wire line
Power
Transmission line
Mechanical filter
88, 225
38
35
38
38
-
204
694
416
731
733
732
734
731
734, 736
74
32
30
37
23
134
107, 112, 218
199, 245
199
731
37
210
80
210
-
-
-
"
SSB
"
Stage
"
Transistor
Triode
"
"
337, 339
336
209
100
211
Tube
79
Mobile, Antenna coupling
"
"
Control circuit
Dynamotor
Equipment construction
Filter circuits
Noise limiter
Noise sources
Power supply
Mode of resonance
516, 520
-
521
522
design -...520, 511
Modulation
" Amplitude
Automatic control
Bandwidth
Cathode
37
Class
35
Constant efficiency
Distortion
Frequency
Grid
Heising
Index, F -M
Pattern, oscilloscope
Percentage
saturation....36
35, 37
36
83
48
93
408, 428
B
Suppressor
Transformer
Variable efficiency
--
Velocity
Modulator
"
740
738
737
731, 733
739
745
"
"
Adjustment
Balanced
S.S.B.
Beam deflection
Bias -shift
Cathode follower
Class B
Construction
Crosby
Diode
Matching
impedance match
731,735
-
740, 744
220
514
512
525
515,517,697
230
237
280, 669
670
281
295
292
284
305
308
250, 286
292
310
178
282
311,315
Phase
Plate
Screen
444
441, 499
438
425
752
13, 413
36
64
571
42
24
802
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249, 291, 293
286
290
293
283
81
676
328, 334, 339
350
305
288
125
672
590
328
125, 127
"
Reactance tube
"
"
Semi -conductor
312
104
336
669
683
697
SSB
"
Tetrode
Zero bias
" 200 -watt
Molecule
Monitor oscilloscope
Morse code
"
21
179
15
Mu factor (µ)
78
Multee antenna
Multi -band antenna
Multiplication, electronic
Multiplier, frequency
Multiplier resistor
Multi -range meters
Multivibrator
circuits
Multivibrator, free running
Multi -wire doublet
Music, def. of
Music systems
scale
Mutual conductance
Mutual coupling
Mutual coupling, antennas
Mutual inductance
Mycalex, dielectric
-
436
432, 510
198
256
732
732
102, 188
189
Ohm's Law (complex quantities)
Ohm's law for magnetic circuit
Ohm's Law (Resonant Circuit)
Ohms - per -volt
Omega, def. of
One -shot multivibrator
On -off circuits
Open wire line
Operating desk, constuction of
Operational amplifier
Optimum working frequency
Orbital electron
Oscillation, parasitic
Oscillator
Beat
"
"
425, 439
134
138, 135
73
-
Procedure
R
-F
Shunt
Test
Octave, def. of
Oersted, def. of
Ohm
Ohm's Law
Ohmmeter, low range
Ohm's Law (a.c.)
-
241
173
245
238
247
245
192
247
670
Pierce
245
199
263
250
107
RC
191
Relaxation
T.p.t.g.
Transistor
Transistron
V.F.O.
102
239
251
615
277
274
252
276
100
-
19
242
240
239
NBS bridge -T
Overtone
Phase shift
253
Generator, silicon
Limiters
Mobile
Mixer
Regulator
sources
Suppression
Thermal agitation
Voltage
Wow and flutter
Nomenclature, color code
Nomenclature, component
Nonconductor
Nondirectional VHF antenna
Non -linear distortion
Non-linear function
Non -resonant transmission line
Nonsinusoidal wave
N -P -N transistor
"Nuvistor" tube, use
21
277, 361, 608, 615
222
102, 189, 190
311
192
141
Transistor
-
199
414
239
Miller
Neutralizing procedure
NI (ampere turns)
Noise
Factor
" Check, mobile
-
Franklin
Free running
Harmonic
Hartley
Keying
471
37
32
417
724
192
248
E.c.o.
216
N
Narrow -band F -M
NBS bridge -T oscillator
Negative feedback loop
Negative peak clipping
Network, R -L integrator
Networks, L and Pi
Neutralization
Amplifier
Capacitance
" Grid
" Grounded -grid
-
Blocking
Bridge -T
Circuits
Clapp
Colpitts
Code practice
Crystal
Dynatron
49
36
54
732
44
189
195
62
152
524
749
225, 512
210
524
225, 523
191
101
241
242
Wein- bridge
191
Oscilloscope
170
" Circuit
174
" Linearity tracer
183
" Lissajous figures
176
Modulation pattern -phase patterns.178, 177
1.
Monitor
179, 181, 750
Sideband measurements
182
Trapezoidal pattern -wave pattern..178, 180
peak
Output,
power
124
Overloading, TV receiver
367
Overtone crystals
247
Overtone, music
135
Oxide filament
67, 69
188
151
-
135
P
Paper dielectric
Parallel circuit
resistance
" Diode limiter
527
526
22
478
135
202
417
Feed
271
"
Resonance
54
745
Wire line measurements
Parasitic antenna, Design chart
" Antenna, VHF
"
547
"
Beam design
Check for
"
Coupling
Element, antenna
o
Oscillation
135
36
Resonance
band, I -F
band, mechanical filter
band, SSB filters
Patterns, antenna
Pass
Pass
Pass
23, 24
733
45
25
186
"
"
58
92
32
803
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494
484
490
364
360
493
277, 361, 608
360
217
221
332
400, 456
PE -103
Peak
Peak
Peak
Peak
Peak
Peak
522
dynamotor
amplifier current
current (a.c.)
1
envelope power,
limiter
SSB
noise limiter
power output
Peaked wave
Pentode amplifier
Pentode tube
Period, sound
Permeability
Phantom signal
angle (a.c.)
Phase
" Angle
difference
" Distortion
" Inverter
Modulation
Shift, clipping
Shift, feedback
Shift, oscillator
Phasing, antenna
Phasing generator, SSB
Phasing networks, audio
Phonograph reproduction
Photoelectric emission
- -
Pickups
Pickup, spurious
Pierce oscillator
-
Pi- network amplifier
design
Pi- network chart
Pi- network coupling
Pilot carrier, SSB
Pitch, sound
Plate current flow
Plate current, static
Plate detector
Plate efficiency amplifier
Plate modulation
Plate
P -N -P
Point
Polar
resistance
transistor
45
324
185
225
124
59
120
77
134
36
378
46
177, 178
109, 135
115
311, 315
299
193
191
462
333
334
136
67
137
380
245
609
265, 263
449
323
134
257
166
222
159
249, 291
73, 74
91
contact transistor
-
notation
Polarization, antenna
VHF
Polyphase rectifier
Potential difference
Potential, electrode
Potential energy
Power amplifier design
Power amplifier triode
Power, feed through
Power, gain
Power gain antenna
Power gain SSB
Power measurement
Power interlock
Power -line filter
Power, resistive
Power supplies
" A.C. -D.C.
Bridge
Design
Dual voltage
Mobile
Oscilloscope
Regulated
Screen
Three -phase
1.
25
Transistorized
Voltage doubler
Voltage multiplier
Voltage quadrupler
92
48
400, 475
693
22
106
30
602
118
612
94
401
324
731, 735
"
61
133
235, 348
409 -416
414
Propagation
Propagation, sporadic -E
Pulsating alternating current
Pulse- repetition frequency
Push -pull amplifier
Push -pull transformer
Push -pull tripler
doubler
Push -to -talk circuit
45
103, 190
604, 121
-
113
258
522
Q
158
405
356
Q amplifier tank
Q antenna
Q coils
55
Q, def. of
Q loaded
57
Q multiplier
Q tank circuit
Q transformer
Q tuned circuit
234
259
425
214
Quad antenna
Quadrant, sine
Quantity, complex
Quench oscillator
467
43
49
207
R
43
Radian, def. of
Radiation, angle of
Radiation, def. of
Radiation, harmonic
Radiation pattern, distortion
Radiation resistance
Radiator, isotropic
Radiator cross- section, VHF
Radio frequency, def. of
Radio mathematics
Radio propagation
Radio teletype
Ramp function
Ratio detector
Ratio, image
amplifier
RC
audio network
391
RC
differentialor
225
RC
oscillator
29
RC
time constant
684
694
716
713
716, 718
697
173
712
267
RC
703
695
695
695
138
137
212
Preselector
Primary transformer
Probe, R.F. (v.t.v.m.)
Product detector
RC
517
716
717
718
383
709
99
386
300 volt
1500 volt
" 2500 volt
Power system, primary
Power transformer
Power transistor
Powerstat auto- transformer
Preamplifier, hi -fi
Pre -emphasis, recording
"
--
407, 456, 474
399
369
474
400, 491
406
475
42
752
409
322
203
319
211
integrator
circuits
transient
Reactance, antenna
Reactance, capacitive
Reactance, leakage
-
191
inductive
Reactance, resonance
Reactance tube modulator
"
Broadcast transistorized
"
DX
operator
Mobile transceiver, 10
804
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-
60
38
net
404
46
63
47
312
201
Read -out
Receiver, alignment
Receiver, superheterodyne
Receiver, transistor
Receivers, High -frequency
"
109
139
59
m.
180, 233
208
103, 529
526
529
564
555
"
"
Transceiver, 10 -15 m.
Transistorized b.c.
539
529
229
317
Receivers, UHF
Reception,
Reception,
Reception,
Recording,
Recording,
Recording,
Recording,
frequency modulation
mobile
347
Rectification, audio
Rectification, stray
"
Hash
"
Mercury vapor
Selenium
Silicon
Type meter
"
"
"
"
"
"
-
polyphase
Vacuum
Voltage doubler
Voltage quadrupler
V.t.v.m.
Reflected impedance
Reflected voltage, transmission line
Reflection, ionospheric
Ref lectometer, coaxial
Reflectometer, SWR meter
Regeneration
Regulated power supply
Regulation, power line
Regulation, power supply
Regulation, voltage
Regulator noise
Regulator tube
Regulator tube (VR)
Regulator, voltage
Rejection notch, I -F
Rel, def. of
Relaxation oscillator
Reluctance, def. of
Remote cut -off tube
Residual magnetism
1.
Capacitance filter
Dynamic
Gain
Ground
Input
Load
Plate
-
-
parallel
Filter
Impedance
Mode
feedback
ground
R -F shielding
Rhombic antenna
Rhumbatron cavity
RIAA equalizer curve
Ribbon TV line
"Ribbon" (TV) transmission line
Ring diode modulator
Ripple factor, filter
Ripple voltage
RL circuit
R -L integrator network
RL transient
RLC circuits
Root- mean -square (a.c.)
Rotary beam antenna
Rotator, antenna
Ruggedized tube
R -F
78
-Q
linear amplifier
Safety bleeder
S- curve,
406
107, 214
series
25
119
73, 74
400
29
23
27, 709
352
527
34
series
353
732
661
24
47
400
56
53, 54
688
54
230
61 2
611, 612
609
607, 608
608
609
604
608
606
270, 357
736
272
358
358
460, 482
230
137
474
418
329
688
686
40
199
38
48
45
490
508
88
S
37
95
94
-
chokes
dummy loads
R -F
36
102, 188
36
Resistor multiplier
Resistor, non -inductive, use of
Resistor, typical
Resonance
Antenna
Current
-F
R -F
686
524
686
709
687
220
Radiation
Resistive power
Resistivity, table of
Resistor, bleeder
Resistor, characteristics of
Resistor, color code
Resistor, equalizing
Resistor, inductance of
Curve
R
384
715
-
608
Grounded -grid
Grounded screen
Inductive tuning
Instability
neutralization
Oscillation
Pi- network
Push -pull
Self -neutralization
Tetrode
-
1.
687
parallel
166
168
B
Construction
690
689
689
694
693, 694
695
696
734
693
695
695
133
57, 63
743
414
742
742
206
23
Internal
"
Class
71 2
Resistance
"
R -F
45
Rectifier, bridge
"
Full -wave
" Half -wave
"
alignment
amplifier
" Class AB'
R -F
379
379
Rectified A.C.
360
140
230
53, 54
420
112, 136
172
234
211, 249
Response, audio
Return trace
constant amplitude, velocity ....136, 137
136
crossover point
136
high fidelity
pre- emphasis
137
RIAA curve
-
181
"
Oscilloscope pattern
Parasitic
" Speaker
Resonant cavity
Resonant circuit
Resonant transmission line
511
SSB
"
Safety precautions
Saturation current
Saturation, magnetic
Sawtooth wave
Scalar notation
Scale, musical
Scatter, ionospheric
Scatter signals
Screen, CRT
Screen grid keying
Screen grid tube
Screen lead inductance
Screen modulation
Screen supply
Secondary emission
Secondary transformer
Selective fading, SSB
Selectivity, arithmetical
Selectivity chart
Selectivity control, I -F
Selectivity, mobile reception
Selectivity, resonant circuit
Selectivity, tuned circuits
Selenium rectifier
Self inductance
805
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166
391
389
72
36
59, 172
48
135
414
415
87
395
77
255
286
267
71, 77
61
327
209
338
219
513
54
339
695
37
Self neutralization amplifier
Selsyn indicator
Semi -conductor
Semi -conductor, heat sink
Semi- conductor rectifier
Semi -conductor, zener
Series -cathode modulator
Series circuit
Series -derived filter
Series -diode limiter
Series feed
Series -feed amplifier
Series -parallel circuit
resonance
Series resistance
-
Shape factor, -F
Shielding, R -F
Shot effect
Shunt -derived filter
Shunt loading, a.v.c.
Shunt, meter
Shunt neutralization
Shunt -regulated bias supply
Sidebands, def. of
Signal, error
Signal, phantom
SSB
Signal -to- distortion ratio
Signal -to -noise ratio
Silicon crystal noise generator
Silicon rectifier
Sine wave
Single -ended amplifier
Single sideband (S513)
" Envelopes
" Filter
" Jr. exciter
" Measurements
" Reception
" Transmission
Single signal reception
Single swing oscillator
Single -wire antenna tuner
Single -wire feed, antenna
Single -wire feeder
Six- shooter antenna
Skeleton VHF antenna
Skin effect
Skip distance
Sky wave
Sleeve antenna
Slide -wire bridge
Slope detection
Slotted transmission line
S -meter circuits
Soldering techniques
Sound in air
Space charge
Space wave
Speaker response
Speaker tweeter
Specific resistance
Speech amplifier construction
I
-
Speech
Speech
Speech
Speech
Speech
clipper, circuitry
clipper filter
clipping
filter, high level
waveforms
608
510
90, 21
104
695
105
295
24
64
185
271
604
25
25, 53
217
358
153
64
224
731
252
712
280
193
378
153, 340
152
749
696
42, 43
603
283, 323, 325
326
324
342
150
347
323
220
189, 190
452
437
426
468
-
Standard
Standard
Standard
Standing
Standing
frequency
frequency, WWV
208
740
224
727
134
71, 78
410
140
139
23
677
678
302
124, 298, 670
741
525
166
524
Static, wheel
Steering diode
Step -by -step counter
Sterba antenna
Storage time, diode
Storm, ionospheric
Stray capacitance
103
190
465
103
"Strip line" amplifier
Stub match, antenna
Summation amplifier
Summation voltage
Summing amplifier
Sunspot cycle
Superheterodyne receiver
Super- regenerative detector
Suppression, audio
Suppression circuits, parasitics
Suppressor
Suppressor modulation
Suppressor, splatter
Surface wave
Surge impedance
Susceptance, def. of
Sweep frequency
Switching, transistor action
bridge
meter, antennascope
meter, balanced line
meter, bridge
meter, reflectometer
meter, "twin lamp"
Symmetry, amplifier
Synchronization, generator
Synchronizing voltage
Synchronous detection,
55
414
410
476
738
399, 419
Static, corona
Static plate current
SWR
SWR
SWR
SWR
SWR
SWR
-
DSB
806
www.americanradiohistory.com
415
216
595
499
200
197
201
414, 415
208
207
304
362
77
290
300
410
420
52
172
704
454, 746
747
745
741
742
745
605
173
172
349
T
capacitance
circuit
efficiency
circuit impedance
loading
circuit Q
Technician class amateur license
Teletype, radio
Television interference
Temperature coefficient
Ten -A, SSB exciter
Terman- Woodyard amplifier
Termination, transmission line
Termination, VHF rhombic
Test equipment
Tetrode, grounded grid use
Tetrode modulator
Tetrode, zero -bias
Tetrode neutralization
Tetrode r -f amplifier
Tetrode tube
Thermal agitation noise
Thermionic emission
Thermocouple meter
Three -phase power supply
Threshold voltage
Thyratron tube
Time -base generator
Tank
Tank
Tank
Tank
750
494, 464
739
739
135
pitch
wave
wave indicator
477
303
283, 292, 299
302, 669
Splatter suppressor
414
Sporadic -E propagation
Spotter, frequency
740
Spurious emission
spurious pickup
379, 380
Spurious products, mixer
339
Square wave
square wave test
58, 61, 62
-
-
monitor oscilloscope
stacked dipole
Stacked antenna
SSB,
-
259
55, 57, 58
260, 262
156
12
322
367
32
343
296
421
483
731
614
669
683
254
606
77
188
67
734, 736
517
696
87
171
Time constant
Time sequence keying
T -match antenna
38, 60
396
497
limiter
Tools, radio
T.P.T.G. oscillator
228
TNS
-
Transceivers, High Frequency
" 10 -15 m.
" 10 m. mobile
Transconductance
Transducer
pickup
Transformation, impedance
-
Transformation ratio
Transformer
ampere -turns
" Antenna match
" Auto
" Coupling
"
"
I
-F
-
"
Linear matching
Matching
impedance match
"
Modulator
" Power
Transformerless power supply
Transient circuit (a.c.)
Transient distortion
Transient, RC, RL
transient wave
Transistor
-
"
Bias
Code oscillator
Complementary circuit
Drift
Equivalent circuit
Heat sink
Impedance
Junction
Mixer
Multivibrator
Oscillator
"
"
"
Point -contact
Power
Receiver
Switching action
6 m, transmitter
Transistorized broadcast receiver
Transistorized mobile supply
Transistorized modulator
Transit time
Transitron oscillator
Transmission line
antennascope
" Balanced, SWR meter
Characteristics
Chart
Circuits
Coaxial
Discontinuities
Impedance
Incident voltage
Measurements
Reflected voltage
Resonant
non- resonant
Slotted
Termination
-
-
"
VHF
-
Kilowatt
721
cathode ray
Tracking capacitor
Transceiver
Trace
Transmitter, High- Frequency
" Deluxe 200 -watt
" 813 grounded -grid amplifier
"
350 -Watt grounded -grid
Transmitter
Control circuits
" Design
" Ground
" Keying
" Low Power
" Oscilloscope, monitor of
4 -400A
"Strip- line" amplifier
Transistorized, 6 m.
"Tri- band" amplifier
3 -1000Z amplifier
4CX1000A amplifier
4 -400A amplifier
Trapezoidal pattern, oscilloscope
Trap -type antenna
Travelling wave tube
Travis discriminator
Triode amplifier
Triode mixer
Triode power amplifier
Triode tube
Tripler, push -pull
Triplex antenna
Tropospheric propagation
T- section filter
Tube, vacuum
Tube, VHF design
Tubular transmission line
Tuned circuit, coaxial
Tuned circuit, I -F
Tuned circuit, r.f. amplifier
Tuned circuit Q
Tuned circuits, selectivity
Tuner, antenna
Tuning, bandspread
Tuning indicators
Tuning, inductive
Turnstile antenna
135
38, 58
90, 92
96
19
97
93
95
104, 707
97
TVI
91
filters
TVI -proof enclosures
TVI suppression
TVI -type filter
100
102
101
-
openings
"Twin lamp" SWR meter
Two -band Marconi antenna
529
704
577
529
581
627
617
649
634
595
577
622
661
654
643
178
510
84
318
110
211
118
72
258
471
411
64
67
232
418
230
216
153
214
339
452
215
224
609
476
369, 374
724, 725
371
Tweeter speaker
95
99
65
139
745
434
u
Ultra- linear amplifier
Unidirectional antenna
Uni -potential cathode
Unity coupling
Unloaded Q
703
104
81
241
747
745
418
418
229
419
amplifier
KW -2 grounded -grid amplifier
239
86, 172
215
528
526
539
555
74, 79
137, 138
63
62
61, 62
498
63
113
209
442
425, 63
293
709
694
59
576
142
500
70
269
57
V
Vacuum capacitor
Vacuum tube
"
Amplification
73, 74
Beam power
78
84
107, 87
73
-
421
417
742
tl
740, 744
743
420, 417
740
421
474
Cathode ray
Classes
classification
Conductance
Constant- current curve
Diode
Equivalent noise resistance
Foreign
Gas
387
is
352
390
391
576
179
356
67
Input loading
Klystron
Load line
Magic eye
Magnetron
Mixer
Operation
807
www.americanradiohistory.com
157
71
153
89
87
152
81
74
88
83
79
75
"
"
Parameters
Pentode
Plate resistance
Polarity reversal
"
Remote cutoff
"
"
Vacuum, Shot effect
" Space charge
" Tetro.Je
"
"
"
"
"
"
Thyrtron
Travelling wave
Triode
Upper frequency
Variable mu (A)
VHF
"
Voltage regulator
" Voltmeter
V- Antenna
Variable -mu (11) tube
Variable reluctance pickup
Variac autotronsformer
Vector, sine -wave
Vehicular noise suppression
Velocity modulation
Vertical antenna
Vertical directivity
VHF
amplifier
"
"
106
77
73, 74
76
78
153
89
77
87
84
72
230
154
80
87
130, 734
458
154
137
386
43
523
81
426
407
212
473
475
474
Antennas
Antenna polarization
" Antenna relay
14
" Bands, characteristics
481
" Corner reflector antenna
473
" Definition of
" Discone antenna -ground -plane antenna -.477
" Helical antenna -Horn antenna -- -.479, 482
484
" Multi- element antenna
478
" Nondirectional antenna
362
" Parasitics
474
radiation pattern
" Radiation angle
482
" Rhombic antenna
487
" Screen antenna
476
" Sleeve antenna
595
" "Strip -line" amplifier
474
" Transmission line
476
" Turnstile antenna
475
" Wavelength table
698
Vibrator, split -reed
112, 128
Video amplifier
22
Volt, def. of
110
Amplifier
Voltage
697
" Avalanche
203
" Break
32
" Breakdown
40
" Decay
26, 52
" Divider
117
" Divider, phase inverter
695
" Doubler supply
28
" Drop, summation
39
Gradient
742
Incident
211
Injection
203
Initial condition
44
Instantaneous
693
Inverse
Measurement
731, 733
Multiplier power supply
695
151
Noise
685
Output
Quadrupler supply
695
Reflected
742
Regulation
686, 687
Regulator tube
87, 686, 709
Resonant
54
-
-
"
686
"
197
172
Ripple
Summation
" Synchronizing
" Threshold
Voltmeter
Voltmeter, A -C V -T
Voltmeter, Diode
Voltmeter, vacuum tube
Volt- ohmmeters
Volume compression
696
732
734
734
130, 734
732
670
w
Wagner ground
Watt, def. of
Wave
Carrier
" Ground
-
"
"
"
"
"
"
739
29
205
- --
410
Harmonic
nonsinusoidal
Pattern, oscilloscope
Peaked
sawtooth
58
180
59
Sky
space
Square
transient
Surface
410
58
410
59, 175
283, 299, 670
Waveform
Waveform, speech
Wavelength, def. of
Wavelength table, VHF
Wave- shaping circuits
401
475
185
24
Weston cell
Wheatstone bridge
Wheel static
Wien- bridge oscillator
Williamson amplifier
Wire, litz
Wiring hints
Wiring technique, audio
Workshop practice
layout
W8JK antenna
WWV Transmissions
-
738
524
191
141
55
527
142
720, 729
469
739
x
X -array
antenna
465
Y
Y, (admittance, def.
-
of)
52
Yogi, adjustment
Yagi antenna
Feed systems
Yogi antenna, VHF
Yogi, constructions
Yoke match
505
494
484
502
494
z
Zener diode
Zeppelin antenna
Zero -bias modulator
Zero -bias tetrode
808
www.americanradiohistory.com
105
423
683
683
CONVERT SURPLUS RADIO GEAR
INTO USEFUL EQUIPMENT
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volumes give complete conversion data,
including instructions, photos and diagrams
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$3.00 ea. (foreign $3.50)
3 Volumes
RADIO CONVERSION MANUALS
Contain data which has become standard for the most commonly available surplus items. Each conversion shown
proved by testing.
yields a practical piece of equipment
ARC -5; BC -221, 224, 312. 342, 348, 412, 453/455, 457/459, 624/625, 645, 696, 946B, 1068A/
VOLUME I
1161A; PE -103A; SCR -211, 268/271, 274N, 522. 542. TBY; Electronic Surplus Index; Cross Index of A/N V.T.
and Commercial Tubes.
AIC; AM -26; APS -13; ARB (Schematic only); ARC -5; ART -13; ATC; AVT -112A; BC -I91, 357, 375.
VOLUME II
454-455, 457/459, 946B, 1206; GO -9; LM; R -26 -27 /ARC -5; R -28 /ARC -5; SCR -274N; TA- 128/12C; TBW;
T -23 /ARC -5; Selenium- Rectifier Power Units; Simplified Coil- Winding Data; Surplus Beam Rotating Mechanisms.
APN -1; ARC -5; ART -13; BC -191, 312. 342, 348, 375, 442, 453, 455, 456 to 459, 603, 624,
VOLUME III
696, 1066, 1253; CBY -52000 series; COL- 43065; CRC -7; DM-34D; DY -8 or DY -2A /ARR -2; FT -241A; LM; MBF;
MD7 /ARC -5; RM -52, 53: R -9 /APN -4; R-28 /ARC -5; RT -19 /ARC -4; RT -159; SCR -274N series; SCR -508. 522,
528, 538; T -15 to T -23 /ARC -5; URC -4; WE- 701 -A. Schematics only: APA -10; APT -2; APT -5; ARR -2; ASB -5;
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APN -1; APS -13; ARB; ARC -4; ARC -5 (L.F.);
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VOLUME I
ARC -5 (V.H.F.); ARN -5; ARR -2; ASB -7; BC -222, 312, 314, 342, 344, 348. 603, 611, 624. 652, 654, 659. 669,
683, 728, 745, 764. 779, 794, 923, 1000, 1004, 1066, 1206, 1306, 1335; BC -AR -231; CRC -7; DAK -3; GF -11;
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SURPLUS
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EDITORS and ENGINEERS, Ltd.
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TELEPHONE
LICENSE
MAC
PREPARES YOU FOR THE F.C.C.
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Now, in one convenient volume, complete study -guide questions with clear, concise answers for preparation for all U.S.A.
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CONTAINS FOUR ELEMENTS:
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EDITORS and ENGINEERS, Ltd. Summerland California
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440 Pout Ave So.
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'TUBE
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IN 3 MANUALS
WORLD'S RADIO TUBES
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Characteristics of all existing radio tubes
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All replacement tubes for a given type, both
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