Radio Handbook 16 1962

User Manual: Radio-Handbook-16-1962

Open the PDF directly: View PDF PDF.
Page Count: 811

DownloadRadio-Handbook-16-1962
Open PDF In BrowserView PDF
1.1161ffliale:n0

This book is revised and brought up
to date (at irregular intervals) os
necessitated by technical progress.

111E
11111110
II

1111ß001i
Sixteenth Edition

WILLIAM

I. ORR, W6SAI

Editor, 16th Edition

The Standard of the Field

for

-

advanced amateurs
practical radiomen
practical engineers

practical technicians

Published and distributed to the electronics trade by

EDITORS and ENGINEERS, Ltd.
Dealers: Electronic distributors, order from us. Bookstores,
Taylor, Hillside, N.J. Export (exc. Canada). order from N.M.

Summerland

www.americanradiohistory.com

,

California

newsdealers order from Baker li
Snyder Co., 440 Park Ave. So., N.Y. 16.

libraria.

THE

HANDBOOK

RADIO
SIXTEENTH

EDITION

Copyright, 1962, by

Editors and Engineers, Ltd.
Summerland, California, U.S.A.

Copyright under Pan -American Convention
All Translation Rights Reserved

Printed in U.S.A.

The "Radio Handbook" is also available on special order in Spanish and
Italian editions; French, German, and Flemish -Dutch editions are in
preparation or planned.
Outside North America, if more convenient, write: (Spanish) Marcombo, S.A., Av.
Jose Antonio, 584, Barcelona, Spain; (Italian) Edizione C.E.L.I., Via Gandino 1,
Bologna, Italy; (French, German, Flemish- Dutch) P. H. Brans, Ltd., 28 Prins Leopold
St., Borgerhout, Antwerp, Belgium.

Other Outstanding Books from the Same Publisher
(See Announcements at Back of Book)
THE RADIOTELEPHONE LICENSE MANUAL

THE SURPLUS RADIO CONVERSION MANUALS
THE SURPLUS HANDBOOK
THE WORLD'S RADIO TUBES

(

RADIO TUBE VADE MECUM)

TILE WORLD'S EQUIVALENT TUBES ( EQUIVALENT TUBE VADE MECUM)

THE WORLD'S TELEVISION TUBES (TELEVISION TUBE VADE MECUM)

www.americanradiohistory.com

THE RADIO

HANDBOOK

16th Edition

Table of Contents
Chapter One. INTRODUCTION TO RADIO
Amateur Radio
-1
Station and Operator Li
-2
The Amateur Bands
-3
Starting Your Study
-4

11

Chapter Two. DIRECT CURRENT CIRCUITS
The Atom
2 -1
Fundamental Electrical Units and Relationships
2 -2
Capacitors
Electrostatics
2 -3
Magnetism and Electromagnetism
2 -4
RC and RL Transients
2 -5

21

Chapter Three. ALTERNATING CURRENT CIRCUITS
Alternating Current
3 -1

41
41

1

1

1

1

-

3 -2
3 -3

3 -4
3 -5

11

12

12
14

21

22
30
35
38

Resonant Circuits
Nonsinusoidal Waves and Transients

53

Transformers
Electric Filters

61

63

Chapter Four. VACUUM TUBE PRINCIPLES
Thermionic Emission
4 -1
4 -2
The Diode
4 -3
The Triode
4 -4
Tetrode or Screen Grid Tubes
Mixer and Converter Tubes
4 -5
Electron Tubes at Very High Frequencies
4 -6
4 -7
Special Microwave Electron Tubes
The Cathode -Ray Tube
4 -8
4 -9

Gas Tubes

4 -10

Miscellaneous Tube Types

58

67
67
71

72

77
79
80
81

84
87
88

Chapter Five. TRANSISTORS AND SEMI -CONDUCTORS
Atomic Structure of Germanium and Silicon
5 -1
Mechanism of Conduction
5 -2
The Transistor
5 -3
Transistor Characteristics
5 -4
Transistor Circuitry
5 -5
Transistor Circuits
5 -6

3

90
90
90
92
94
96
103

Chapter Six. VACUUM TUBE AMPLIFIERS
6 -1

Vacuum Tube Parameters

6 -2

Classes and Types of Vacuum -Tube Amplifiers

6 -3

6 -8

Biasing Methods
Distortion in Amplifiers
Resistance- Capacitance Coupled Audio- Frequency Amplifiers
Video -Frequency Amplifiers
Other Interstage Coupling Methods
Phase Inverters

6 -9

D -C

6 -10

Single -ended Triode Amplifiers
Single -ended Pentode Amplifiers
Push -Pull Audio Amplifiers
Class B Audio Frequency Power Amplifiers
Cathode- Follower Power Amplifiers
Feedback Amplifiers
Vacuum -Tube Voltmeters

6 -4
6 -5

6 -6
6 -7

6 -11
6 -12
6 -13
6 -14
6 -15
6 -16

Amplifiers

106
106
107
108
109
109
113
113
115
117
118
120
121

123
127
129
130

Chapter Seven. HIGH FIDELITY TECHNIQUES
7 -1
The Nature of Sound
7 -2
The Phonograph
7 -3
The High Fidelity Amplifier
7 -4
Amplifier Construction
7 -5
The "Baby Hi Fi"
7 -6
A Transformerless 25 Watt Music Amplifier

134
134
136
138
142
143
146

Chapter Eight. RADIO FREQUENCY VACUUM TUBE AMPLIFIERS
Tuned RF Vacuum Tube Amplifiers
8 -1
Grid Circuit Considerations
8 -2
Plate- Circuit Considerations
Radio- Frequency Power Amplifiers
8 -3
Class C. R -F Power Amplifiers
8 -4
Class B Radio Frequency Power Amplifiers
8 -5
Special R -F Power Amplifier Circuits
8 -6
Class ABI Radio Frequency Power Amplifiers

151
151
151

153
154
154
159
162

166

Chapter Nine. THE OSCILLOSCOPE
9 -1
A Typical Cathode -Ray Oscilloscope
Display of Waveforms
9 -2
9 -3
Lissajous Figures
9 -4
Monitoring Transmitter Performance with the Oscilloscope
9 -5
Receiver I -F Alignment with an Oscilloscope
9 -6
Single Sideband Applications

170

Chapter Ten. SPECIAL VACUUM TUBE CIRCUITS
10 -1
Limiting Circuits
10 -2
Clamping Circuits
10 -3
Multivibrators
10 -4
The Blocking Oscillator
10 -5
Counting Circuits
10 -6
Resistance - Capacity Oscillators
10 -7
Feedback

185
185
187
188
190
190

4

www.americanradiohistory.com

170
175
176
179
180
182

191

192

194

Chapter Eleven. ELECTRONIC COMPUTERS
Digital Computers
11 -1
Binary Notation
11 -2
Analog Computers
11 -3
-4
-5
11 -6
11 -7
11
11

195
195
197
199

The Operational Amplifier
Solving Analog Problems
Non -linear Functions
Digital Circuitry

200
202
204

Chapter Twelve. RADIO RECEIVER FUNDAMENTALS
Detection or Demodulation
12 -1
Superregenerative Receivers
12 -2
Superheterodyne Receivers
12 -3
Mixer Noise and Images
12 -4

211

Stages

12 -5

R -F

12 -6

Signal- Frequency Tuned Circuits
I -F Tuned Circuits
Detector, Audio, and Control Circuits

12 -7

12 -8
12 -9

12 -10
12 -11
12 -12

Noise Suppression
Special Considerations in
Receiver Adjustment
Receiving Accessories

U -H -F

Receiver Design

Chapter Thirteen. GENERATION OF RADIO FREQUENCY ENERGY
Self -Controlled Oscillators
13 -1
Quartz Crystal Oscillators
13 -2
Crystal Oscillator Circuits
13 -3
Radio Frequency Amplifiers
13 -4
Neutralization of R.F. Amplifiers
13 -5
13 -6
13 -7
13 -8
13 -9
13 -10
13 -11
13 -12
13 -13
13 -14
13 -15

Neutralizing Procedure
Grounded Grid Amplifiers
Frequency Multipliers
Tank Circuit Capacitances
L and Pi Matching Networks

Grid Bias
Protective Circuits for Tetrode Transmitting Tubes
Interstage Coupling
Radio- Frequency Chokes
Parallel and Push -Pull Tube Circuits

Chapter Fourteen.
14 -1
14 -2
14 -3

R -F

FEEDBACK

Feedback Circuits
Feedback and Neutralization of a Two -Stage R -F Amplifier
Neutralization Procedure in Feedback -Type Amplifiers
R-F

Chapter Fifteen. AMPLITUDE MODULATION
.. ..
Sidebands ..
15 -1
Mechanics of Modulation
15 -2
Systems of Amplitude Modulation
15 -3
15 -4

15 -5

205
205
207
208
210

Input Modulation Systems
Cathode Modulation

5

www.americanradiohistory.com

214
216
223
225
229
233
234

237
237
242
245
249
250
253
256
256
259
263
265
267
268
270
271

272
272
275
277

280
280
281

283
290
295

15 -6

-7
15 -8
15

The Doherty and the Terman-

Woodyard Modulated Amplifiers 296
298
The Bias -Shift Heising Modulator
305
Speech Clipping

Chapter Sixteen. FREQUENCY MODULATION AND RADIOTELETYPE
TRANSMISSION
16 -1
Frequency Modulation
16 -2
Direct FM Circuits
16 -3
Phase Modulation
16 -4
Reception of FM Signals
16 -5
Radio Teletype

Chapter Seventeen. SIDEBAND TRANSMISSION
17 -1
Commercial Applications of SSB
17 -2
Derivation of Single -Sideband Signals
17 -3
Carrier Elimination Circuits
17 -4
Generation of Single -Sideband Signals
17 -5
Single Sideband Frequency Conversion Systems
17 -6
Distortion Products Due to Nonlinearity of R-F Amplifiers
17 -7
Sideband Exciters
17 -8
Reception of Single Sideband Signals
17 -9
Double Sideband Transmission
17 -10 The Beam Deflection Modulator
Chapter Eighteen. TRANSMITTER DESIGN
18 -1

Resistors

18 -2

Capacitors
Wire and Inductors
Grounds
Holes, Leads and Shafts
Parasitic Resonances
Parasitic Oscillation in R-F Amplifiers
Elimination of V -H -F Parasitic Oscillations
Checking for Parasitic Oscillations

18 -3
18 -4

18 -5
18 -6

18 -7
18 -8

18 -9

Chapter Nineteen. TELEVISION AND BROADCAST INTERFERENCE
19 -1
Types of Television Interference
19 -2
Harmonic Radiation
19 -3
19 -4

19 -5

Low-Pass Filters
Broadcast Interference
HI -FI Interference

Chapter Twenty. TRANSMITTER KEYING AND CONTROL
20-1
Power Systems
20 -2
Transmitter Control Methods
20 -3
Safety Precautions
20 -4
Transmitter Keying
20 -5
Cathode Keying
20 -6
Grid Circuit Keying
20 -7
Screen Grid Keying
20 -8
Differential Keying Circuits

6

www.americanradiohistory.com

308
308
311

315
317

322

323
323
324
328
330
336
340
342
347
349
350
352
352
354
356
358
358
360
361

362
364

367
367
369
372
375
382
383
383
387
389
391
393
394
395
396

Chapter Twenty -One. RADIATION, PROPAGATION AND TRANSMISSION
LINES
21 -1
21

-2

21

-3

399

.._

399

Radiation from an Antenna
General Characteristics of Antennas

.-

400

Radiation Resistance and Feed -Point Impedance
Antenna Directivity

403

409

21 -8

Bandwidth
Propagation of Radio Waves
Ground -Wave Communication
Ionospheric Propagation -_

21 -9

Transmission Lines

416

21 -10

Non -Resonant Transmission Lines

417

21 -11

Tuned or Resonant Lines

420

Line Discontinuities

421

21 -4
21

-5

21

-6

21

-7

21

-12

406

409
410
412

422

Chapter Twenty -Two. ANTENNAS AND ANTENNA MATCHING
End -Fed Half -Wave Horizontal Antennas
22 -1
Center -Fed Half -Wave Horizontal Antennas
22 -2

422
423

__

22 -3

The Half -Wave Vertical Antenna

426

22 -4

The Ground Plane Antenna

427

22 -5

The Marconi

22 -6

Space- Conserving Antennas

430

22 -7

Multi -Band Antennas
Matching Non -Resonant Lines to the Antenna
Antenna Construction
Coupling to the Antenna System

432

Antenna Couplers
A Single -Wire Antenna Tuner

450

22 -8
22 -9
22 -10
22 -11

22 -12

428

Antenna

438
444

447
452

455

Chapter Twenty-Three. HIGH FREQUENCY ANTENNA ARRAYS
Directive Antennas
23 -1
Long Wire Radiators
23 -2

455

457
458
460

23 -3

The V Antenna

23 -4

The Rhombic Antenna

23 -5

Stacked -Dipole Arrays

461

23 -6

Broadside Arrays

464

23 -7

End -Fire Directivity

469

23 -8

Combination End -Fire and Broadside Arrays

471

473
473

Chapter Twenty -Four. V -H -F AND U -H -F ANTENNAS
Antenna Requirements
24 -1
24 -2
Simple Horizontally- Polarized Antennas

475

24 -3

Simple Vertical -Polarized Antennas

24 -4

The Discone Antenna

24 -5

Helical Beam Antennas

476
477
479

24 -6

The Corner -Reflector and Horn -Type Antennas

481

24 -7

VHF

Horizontal Rhombic Antenna
Multi- Element V-H -F Beam Antennas

482

24 -8

_.....

7

www.americanradiohistory.com

484

Chapter Twenty-Five. ROTARY BEAMS
25 -1

Unidirectional Parasitic End -Fire Arrays (Yogi Type)

25 -2

The Two Element

25 -3

The Three -Element

25 -4

Feed Systems

Beam .___

490
490
490
492

_

Array

494

25 -6

for Parasitic (Yogi) Arrays
Unidirectional Driven Arrays
Bi- Directional Rotatable Arrays

25 -7

Construction of Rotatable Arrays

502

25 -8

Tuning the Array

505

25 -9

Antenna Rotation Systems
Indication of Direction
"Three- Band" Beams

509

25 -5

25 -10
25 -11

26 -3
26 -4
26 -5

501

510
510

Chapter Twenty -Six. MOBILE EQUIPMENT DESIGN AND
26 -1
Mobile Reception
26 -2

500

INSTALLATION

Mobile Transmitters
Antennas for Mobile Work
Construction and Installation of Mobile Equipment
Vehicular Noise Suppression

Chapter Twenty-Seven. RECEIVERS AND TRANSCEIVERS
27 -1
Circuitry and Components
27 -2
A Simple Transistorized Portable B -C Receiver
27 -3
27 -4

An Inexpensive Bandpass-Filter Receiver
A Compact Transceiver for 10 and 15 Meters

27 -6

"Siamese" Converter for Six and Two Meters
A Deluxe Mobile Transceiver

27 -7

A Deluxe Receiver for the DX Operator

27 -5

Chapter Twenty- Eight. LOW

POWER

TRANSMITTERS AND EXCITERS

511

511

517
518
520
523
526
529

529
530
539
547
555
564
.... 577

A Transistorized 50 Mc. Transmitter and Power Supply
A Deluxe 200 -Watt Tabletop Transmitter

578

Strip -Line Amplifiers for VHF Circuits
A "9T0" Electronic Key

595
597

Chapter Twenty -Nine. HIGH FREQUENCY POWER AMPLIFIERS

602
602

28 -1
28 -2
28 -3

28 -4

29 -1

Power Amplifier Design

29 -2

Push -Pull Triode

29 -3

Push -Pull Tetrode

29 -4
29 -5
29 -6

29 -7
29 -8
29 -9
29 -10
29 -11
29 -12
29 -13

Amplifiers

Amplifiers
Tetrode Pi- Network Amplifiers
Grounded -Grid Amplifier Design
A 350 Watt P.E.P. Grounded -Grid Amplifier
The "Tri-Bander" Linear Amplifier for 20 -15 -10
An 813 Grounded -Grid Linear Amplifier
The KW -2. An Economy Grounded -Grid Linear Amplifier
A Pi- Network Amplifier for

C -W, A -M, or SSB
Kilowatt Amplifier for Linear or Class C Operation
A 2- Kilowatt P.E.P. All -Band Amplifier
A 3 -1000Z Linear Amplifier

8

www.americanradiohistory.com

581

604
606
609
612

617
622

627
634
643
649
654
661

Chapter Thirty.

SPEECH

AND AMPLITUDE MODULATION EQUIPMENT

669

30 -1

Modulation

669

30 -2

Design of Speech Amplifiers and Modulators

672

Modulator

673

30 -3

General Purpose Triode Class

30 -4

A 10 -Watt Amplifier- Driver

30 -5

A 15 -Watt Clipper- Amplifier

677
678

30 -6

A 200 -Watt 811 -A De -Luxe Modulator

679

30 -7

Zero Bias Tetrode Modulators

683

B

684

Chapter Thirty -One. POWER SUPPLIES
Power Supply Requirements
31 -1

684

-2

Rectification Circuits

689

-3

Standard Power Supply Circuits

690

31 -4

Selenium and Silicon Rectifiers

695

31 -5

100 Watt Mobile Power Supply

31 -6
31 -7

Transistorized Power Supplies
Two Transistorized Mobile Supplies

697
703
706

31 -8

Power Supply Components

31 -9

Special Power Supplies

31 -10

Power Supply Design

31 -11

-_
300 Volt, 50 Ma. Power Supply
1500 Volt, 425 Milliampere Power Supply

716

A Dual Voltage Transmitter Supply

718
718

31
31

31

-12

31

-13
-14

31

707
709
713
.

.

_

A Kilowatt Power Supply

717

720

Chapter Thirty -Two. WORKSHOP PRACTICE

720
723

32 -1

Tools

32 -2

The

32 -3

TVI -Proof Enclosures

724

32 -4

Enclosure Openings

32 -5

Summation of the Problem

725
725

32 -6

Construction Practice

32 -7

Shop Layout

Material

726
729
731

Chapter Thirty- Three. ELECTRONIC TEST EQUIPMENT
Voltage, Current and Power
33 -1
Measurement of Circuit Constants _
33 -2

731

737

33 -3

Measurements with a Bridge

738

33 -4

Frequency Measurements

739

33 -5

Antenna and Transmission Line Measurements

740

33 -6

A Simple Coaxial

742

33 -7

Measurements on Balanced Transmission Lines

33 -8

A

33-9

The Antennascope

33 -10

A Silicon Crystal Noise Generator

747
749

33 -11

A Monitor Scope for AM and SSB

750

"Balanced"

Reflectometer

Chapter Thirty -Four. RADIO

MATHEMATICS AND

744
745

SWR Bridge

CALCULATIONS

9

www.americanradiohistory.com

752

FOREWORD TO THE SIXTEENTH EDITION
Over two decades ago the historic first edition of the RADIO HANDBOOK
was published as a unique, independent, communications manual written
especially for the advanced radio amateur and electronic engineer. Since that early
issue, great pains have been taken to keep each succeeding edition of the RADIO

HANDBOOK abreast of the rapidly expanding field of electronics.
So quickly has the electron invaded our everyday affairs that it is now no
longer possible to segregate one particular branch of electronics and define it as
radio communications; rather, the transfer of intelligence by electrical means
encompasses more than the vacuum tube, the antenna, and the tuning capacitor.

Included in this new, advanced Sixteenth Edition of the RADIO HANDBOOK
are fresh chapters covering electronic computers, r.f. feedback amplifiers, and high
fidelity techniques, plus greatly expanded chapters dealing with semi- conductors
and special vacuum tube circuits. The other chapters of this Handbook have been
thoroughly revised and brought up to date, touching briefly on those aspects in
the industrial and military electronic fields that are of immediate interest to the
electronic engineer and the radio amateur. The construction chapters have been
completely re- edited. All new equipments described therein are of modern
design, free of TV! producing problems and various unwanted parasitic
oscillations.
The writing and preparation of this Handbook would have been impossible
without the lavish help that was tended the editor by fellow amateurs and sympathetic electronic organizations. Their friendly assistance and helpful suggestions
were freely given in the true amateur spirit to help make the 16th edition of the
RADIO HANDBOOK an outstanding success.
The editor and publisher wish to thank these individuals and companies whose
unselfish support made the compilation and publication of this book an interesting and inspired task.
-WILLIAM I. ORR, W6SAI, 3A2AF, Editor
Thomas Consalvi, W3EOZ,
Barker & Williamson, Inc.
Claude E. Doner, W3FAL,
Radio Corporation of
America
John A. Evans, W9HRH,
Potter & Brumfield Co.
Wayne Green, W2NSD,
73 Magazine
Jo Jennings, W6EI,
Jennings Radio Mfg. Co.

E. A. Neal, W4ITC,

General Electric Co.
Harold Vance, K2FF,
Radio Corporation of
America
Blackhawk Engineering Co.
H. E. Blaksley, K7ASK
Byron Hunter, W6VML
Clifford Johnson, WOURQ
Herbert Johnson, W7GRA
Thomas Lamb, K8ERV

James G. Lee, W6VAT
Hugh MacDonald, W6CDT
Otto Miller, K6ENX
Robert Moore, W7JNC
B. A. Ontiveros, W6FFF
(drafting)
A. L. Patrick, W9EHW
Raymond Rinaudo, W6KEV
Robert Sutherland, W6UOV
W. H. Sayer, Jr., WA6BAN
Mel Whiteman, W6BZ

www.americanradiohistory.com

CHAPTER ONE

Introduction to Radio
to the teaching of the principles of equipment
design and signal propagation. It is in response
to requests from schools and agencies of the
Department of Defense, in addition to persistent requests from the amateur radio fraternity,
that coverage of these principles has been expanded.

The field of radio is a division of the much
larger field of electronics. Radio itself is such
a broad study that it is still further broken
down into a number of smaller fields of which
only shortwave or high- frequency radio is covered in this book. Specifically the field of communication on frequencies from 1.8 to 450 megacycles is taken as the subject matter for this
work.

1

The largest group of persons interested in
the subject of high-frequency communication is
the more than 350,000 radio amateurs located
in nearly all countries of the world. Strictly
speaking, a radio amateur is anyone interested
in radio non -commercially, but the term is ordinarily applied only to those hobbyists possessing transmitting equipment and a license from
the government.
It was for the radio amateur, and particularly for the serious and more advanced amateur, that most of the equipment described in
this book was developed. However, in each
equipment group, simple items also are shown
for the student or beginner. The design principles behind the equipment for high- frequency
radio communication are of course the same
whether the equipment is to be used for commercial, military, or amateur purposes, the

principal differences

lying

-1

Amateur Radio

Amateur radio is a fascinating hobby with
many phases. So strong is the fascination offered by this hobby that many executives, engineers, and military and commercial operators
enjoy amateur radio as an avocation even
though they are also engaged in the radio field
commercially. It captures and holds the interest of many people in all walks of life, and in
all countries of the world where amateur activities are permitted by law.
Amateurs have rendered much public service through furnishing communications to and
from the outside world in cases where disaster
has isolated an area by severing all wire com-

munications. Amateurs have a proud record of
heroism and service in such occasion. Many
expeditions to remote places have been kept
in touch with home by communication with amateur stations on the high frequencies. The amateur's fine record of performance with the
"wireless" equipment of World War I has been
surpassed by his outstanding service in World

in construction

practices, and in the tolerances and safety
factors placed upon components.
With the increasing complexity of high-frequency communication, resulting primarily from
increased utilization of the available spectrum, it becomes necessary to delve more deeply into the basic principles underlying radio

War II.

By the time peace came in the Pacific in
the summer of 1945, many thousand amateur
operators were serving in the allied armed
forces. They had supplied the army, navy,
marines, coast guard, merchant marine, civil
service, war plants, and civilian defense organizations with trained personnel for radio,

communication, both from the standpoint of
equipment design and operation and from the
standpoint of signal propagation. Hence, it will
be found that this edition of the RADIO HANDBOOK has been devoted in greater proportion
11

www.americanradiohistory.com

Introduction to Radio

12

radar, wire, and visual communications and
for teaching. Even now, at the time of this
writing, amateurs are being called back into
the expanded defense forces, are returning to
defense plants where their skills are critically
needed, and are being organized into communication units as an adjunct to civil defense
groups.
1

Station and Operator Licenses

-2

Every radio transmitting station in the
United States no matter how low its power
must have a license from the federal government before being operated; some classes of
stations must have a permit from the government even before being constructed. And every
operator of a transmitting station must have
an operator's license before operating a transmitter. There are no exceptions. Similar laws
apply in practically every major country.

There are at present six
classes of amateur operator licenses which have
been authorized by the Federal Communications Commission. These classes differ in

"Classes of Amateur
Operator Li

many

respects, so each will

be

discussed

briefly.
(a) Amateur Extra Class. This class of license is available to any U. S. citizen who at
any time has held for a period of two years or
more a valid amateur license, issued by the
FCC, excluding licenses of the Novice and
Technician Classes. The examination for the
license includes a code test at 20 words per
minute, the usual tests covering basic amateur
practice and general amateur regulations, and
an additional test on advanced amateur practice. All amateur privileges are accorded the
holders of this operator's license.
(b) General Class. This class of amateur
license is equivalent to the old Amateur Class
B license, and accords to the holders all amateur privileges except those which may be set
aside for holders of the Amateur Extra Class
license. This class of amateur operator's license is available to any U. S. citizen. The
examination for the license includes a code
test at 13 words per minute, and the usual examinations covering basic amateur practice
and general amateur regulations.
(c) Conditional Class. This class of amateur license and the privileges accorded by it
are equivalent to the General Class license.
However, the license can be issued only to
those whose residence is more than 125 miles
airline from the nearest location at which FCC
examinations are held at intervals of not more
than three months for the General Class amateur operator license, or to those who for any

THE

RADIO

of several specified reasons are unable to appear for examination.
(d) Technician Class. This is a new class
of license which is available to any citizen of
the United States. The examination is the same
as that for the General Class license, except
that the code test is at a speed of 5 words per
minute. The holder of a Technician class license is accorded all authorized amateur privileges in the amateur frequency bands above
220 megacycles, and in the 50-Mc. band.
(e) Novice (.lass. this is a new class of
license which is available to any U. S. citizen
who has not previously held an amateur license of any class issued by any agency of
the U. S. government, military or civilian. The
examination consists of a code test at a speed
of 5 words per minute, plus an examination on
the rules and regulations essential to beginner's operation, including sufficient elementary radio theory for the understanding of those
rules. The Novice Class of license affords
severely restricted privileges, is valid for only
a period of one year (as contrasted to all other
classes of amateur licenses which run for a
term of five years), and is not renewable.
All Novice and Technician class examinations are given by volunteer examiners, as regular examinations for these two classes are
not given in FCC offices. Amateur radio clubs
in the larger cities have established examin
ing committees to assist would -be amateurs
of the area in obtaining their Novice and Technician licenses.
1

-3

The Amateur Bands

Certain small segments of the radio frequen-

cy spectrum between 1500 kc. and 10,000 .fc.
are reserved for operation of amateur radio
stations. These segments are in general agreement throughout the world, although certain
parts of different amateur bands may be used
for other purposes in various geographic regions. In particular, the 40 -meter amateur band
is used legally (and illegally) for short wave
broadcasting by many countries in Europe,
Africa and Asia. Parts of the 80 -meter band
are used for short distance marine work in Europe, and for broadcasting in South America.
The amateur bands available to American radio amateurs aree

The 160 -meter band is divided into 25- kilocycle
segments on a regional
basis, with day and night power limitations,
and is available for amateur use provided no
interference is caused to the Loran (Long
Range Navigation) stations operating in this
band. This band is least affected by the 11-

160 Meters
(1800 Kc. -2000 Kc.)

www.americanradiohistory.com

Amateur Bands

HANDBOOK
year solar sunspot cycle. The Maximum Usable Frequency (MUF) even during the years
of decreased sunspot activity does not usually
drop below 4 Mc., therefore this band is not
subject to the violent fluctuations found on
the higher frequency bands. DX contacts on
on this band are limited by the ionospheric
absorption of radio signals, which is quite
high. During winter nighttime hours the absorption is often of a low enough value to permit trans -oceanic contacts on this band. On
rare occasions, contacts up to 10,000 miles
have been made. As a usual rule, however,
160 -meter amateur operation is confined to
ground -wave contacts or single -skip contacts
of 1000 miles or less. Popular before World
War II, the 160 -meter band is now only sparsely occupied since many areas of the country
are blanketed by the megawatt pulses of the
Loran chains.
The 80 -meter band is the
most popular amateur
band in the continental
United States for local "rag- chewing" and
traffic nets. During the years of minimum sunspot activity the ionospheric absorption on
this band may be quite low, and long distance
DX contacts are possible during the winter
night hours. Daytime operation, in general, is
limited to contacts of 500 miles or less. During the summer months, local static and high
ionospheric absorption limit long distance contacts on this band. As the sunspot cycle advances and the MUF rises, increased ionospheric absorption will tend to degrade the
long distance possibilities of this band. At
the peak of the sunspot cycle, the 80 -meter
band becomes useful only for short-haul communication.
80 Meters

(3500 Kc. -4000 Kc.)

The 40 -meter band is high
Kc) enough in frequency to be
severely affected by the
11 -year sunspot cycle. During years of minimum solar activity, the MUF may drop below
7 Mc., and the band will become very erratic,
with signals dropping completely out during
the night hours. Ionospheric absorption of signals is not as large a problem on this band as
it is on 80 and 160 meters. As the MUF gradually rises, the skip- distance will increase on
40 meters, especially during the winter months.
At the peak of the solar cycle, the daylight
skip distance on 40 meters will be quite long,
and stations within a distance of 500 miles or
so of each other will not be able to hold communication. DX operation on the 40 -meter band
is considerably hampered by broadcasting stations, propaganda stations, and jamming trans40 Meters

(7000 Kc. -7300

13

mitters. In Europe and Asia the band is in a
chaotic state, and amateur operation in this region is severely hampered.
At the present time,
20 Meters
(14,000 Kc.-14,350 Kc.) the 20 -meter band is
by far the most popular

band for long distance contacts. High enough
in frequency to be almost obliterated at the
bottom of the solar cycle, the band nevertheless provides good DX contacts during years
of minimal sunspot activity. At the present
time, the band is open to almost all parts of
the world at some time during the year. During the summer months, the band is active until the late evening hours, but during the winter months the band is only good for a few
hours during daylight. Extreme DX contacts
are usually erratic, but the 20 -meter band is
the only band available for DX operation the
year around during the bottom of the DX cycle.
As the sunspot count increases and the MUF
rises, the 20 -meter band will become open for
longer hours during the winter. The maximum
skip distance increases, and DX contacts are
possible over paths other than the Great Circle
route. Signals can be heard the "long paths,"
180 degrees opposite to the Great Circle path.
During daylight hours, absorption may become
apparent on the 20 -meter band, and all signals
except very short skip may disappear. On the
other hand, the band will be open for worldwide DX contacts all night long. The 20 -meter
band is very susceptible to "fade- outs"
caused by solar disturbances, and all except
local signals may completely disappear for
periods of a few hours to a day or so.

This is a relatively
new band for radio
amateurs since it has
only been available for amateur operation
since 1952. Not too much is known about the
characteristics of this band, since it has not
been occupied for a full cycle of solar activity. However, it is reasonable to assume that
it will have characteristics similar to both the
20 and 10 -meter amateur bands. It should have
a longer skip distance than 20 meters for a
given time, and sporadic -E (short -skip) should
be apparent during the winter months. During
a period of low sunspot activity, the MUF will
rarely rise as high as 15 meters, so this band
will be "dead" for a large part of the year.
During the next few years, 15 -meter activity
should pick up rapidly, and the band should
support extremely long DX contacts. Activity
on the 15 -meter band is limited in some areas,
15 Meters

(21,000 Kc.- 21,450 Kc.)

www.americanradiohistory.com

14

I

n t r o d u c

t

i

o n

t o

R a d

i

o

since the older model TV receivers have a
21 Mc. i -f channel, which falls directly in the
15 -meter band. The interference problems
brought about by such an unwise choice of
intermediate frequency often restrict operation
on this band by amateur stations unfortunate
enough to be situated near such an obsolete
receiver.
Meters
(28.000 Kc.- 29,700 Kc.)
10-

During the peak of the
sunspot cycle, the 10meter band is without
doubt the most popular
amateur band. The combination of long skip
and low ionospheric absorption make reliable
DX contacts with low powered equipment possible. The great width of the band (1700 kc.)
provides room for a large number of amateurs.
The long skip(1500 miles or so) prevents nearby amateurs from hearing each other, thus
dropping the interference level. During the winter months, sporadic -E (short skip) signals
up to 1200 miles or so will be heard. The 10meter band is poorest in the summer months,
even during a sunspot maximum. Extremely
long daylight skip is common on this band, and
and in years of high MUF the 10 -meter band
will support intercontinental DX contacts during daylight hours.
The second harmonic of stations operating
in the 10 -meter band falls directly into television channel 2, and the higher harmonics of
10 -meter transmitters fall into the higher TV
channels. This harmonic problem seriously
curtailed amateur 10 -meter operation during
the late 40's. However, with the new circuit
techniques and TVI precautionary measures
stressed in this Handbook, 10 -meter operation
should cause little or no interference to nearby television receivers of modern design.

At the peak of the sunspot
cycle, the MUF occasionally rises high enough to permit DX contacts up to 10,000 miles or so on
6 meters. Activity on this band during such a
period is often quite high. Interest in this band
wanes during a period of lesser solar activity,
as contacts, as a rule, are restricted to short skip work. The proximity of the 6-meter band
to television channel 2 often causes interference problems to amateurs located in areas
where channel 2 is active. As the sunspot cycle increases, activity on the 6 -meter band will
increase.
Six Meters
(50 Mc. -54 Mc.)

The V -HF Bands
(Two Meters and "Up ")

v -h -f bands are
the least affected by

The

the vagaries of the
sunspot cycle and the Heaviside layer. Their
predominant use is for reliable communication
over distances of 150 miles or less. These

T H E

R A D

I

O

bands are sparsely occupied in the rural sections of the United States, but are quite heavily congested in the urban areas of high popu-

lation.

In recent years it has been found that v -h -f
signals are propagated by other means than by
line -of-sight transmission. "Scatter signals,"
Aurora reflection, and air -mass boundary bending are responsible for v -h -f communication up
to 1200 miles or so. Weather conditions will
often affect long distance communication on

the 2 -meter band, and all the v -h -f bands are
particularly sensitive to this condition.
The other v -h -f bands have had insufficient
occupancy to provide a clear picture of their
characteristics. In general, they behave much
as does the 2 -meter band, with the weather
effects becoming more pronounced on the higher frequency bands.
1

-4

Starting Your Study

When you start to prepare yourself for
amateur examination you will find that the
cuit diagrams, tube characteristic curves,
formulas appear confusing and difficult of

derstanding. But after

the

cirand
un-

a few study sessions
becomes sufficiently familiar with the
notation of the diagrams and the basic concepts of theory and operation so that the acquisition of further knowledge becomes easier
and even fascinating.
As it takes a considerable time to become
proficient in sending and receiving code, it is
a good idea to intersperse technical study sessions with periods of code practice. Many
short code practice sessions benefit one more
than a small number of longer sessions. Alternating between one study and the other keeps
the student from getting "stale" since each
type of study serves as a sort of respite from
the other.
When you have practiced the code long
enough you will be able to follow the gist of
the slower sending stations. Many stations
send very slowly when working other stations
at great distances. Stations repeat their calls
many times when calling other stations before
contact is established, and one need not have
achieved much code proficiency to make out
their calls and thus determine their location.

one

The Code

The applicant for any class of amateur operator license must be able
to send and receive the Continental Code
(sometimes called the International Morse
Code). The speed required for the sending and
receiving test may be either 5, 13, or 20 words
per minute, depending upon the class of license, assuming an average of five characters
to the word in each case. The sending and re-

www.americanradiohistory.com

Learning the Code

HANDBOOK
A

6
C
D
E

.

=
MI

N

O

P
Q

MO

H
I
,J

K

L
M

MED

IEM

T

7

NEI

EMI

V
W

=El 41

X

MIMEE

Y

MIMED MI

Z

MD

ME Ma

S
U

-.

)

2
3
4

Mo =El

5
6

R

F

G

MD

8
9

GM

.

=..
fm. gm,

OM 4=1

1M

ME, MED 4=1

IMP

0
MEANS ZERO. AND IS WRITTEN IN THIS
WAY TO DISTINGUISH IT FROM THE LETTER 'O''
IT OFTEN IS TRANSMITTED INSTEAD AS ONE
LONG DASH (EQUIVALENT TO 5 DOTS)

0

MI

PERIOD (.)

WAIT SIGN (AS)

COMMA (,)

DOUBLE DASH (BREAK)

INTERROGATION (7)
QUOTATION MARK (")

ERROR (ERASE SIGN)

COLON

(

FRACTION BAR( /)
END OF MESSAGE (AR)

)

SEMICOLON

END OF TRANSMISSION (SK)
INTERNAT. DISTRESS SIG. (SOS)

(I)

PARENTHESIS

15

( I

Figure

.

_
MMD
mo
IMID

e
moo

1

rodio
The Continental (or International Morse) Code is used for substantially all non-automatic and
of SOUND,
communication. DO NOT memorize from the printed page; code is a language
must not be learned visually; learn by listening as explained in the text.

ceiving tests run for five minutes, and one
minute of errorless transmission or reception
must be accomplished within the five -minute
interval.
If the code test is failed, the applicant must
wait at least one month before he may again
appear for another test. Approximately 30% of
amateur applicants fail to pass the test. It
should be expected that nervousness and excitement will at least to some degree temporarily lower the applicant's code ability. The
best prevention against this is to master the
code at a little greater than the required speed
under ordinary conditions. Then if you slow
down a little due to nervousness during a test
the result will not prove fatal.

There is no shortcut to code pro ficiency. To memorize the alphabet entails but a few evenings of diligent application, but considerable
time is required to build up speed. The exact
time required depends upon the individual's
ability and the regularity of practice.
While the speed of learning will naturally
vary greatly with different individuals, about
70 hours of practice (no practice period to be
over 30 minutes) will usually suffice to bring
a speed of about 13 w.p.m.; 16 w.p.m. requires
about 120 hours; 20 w.p.m., 175 hours.
Memorizing
the Code

Since code reading requires that individual
letters be recognized instantly, any memoriz-

ing scheme which depends upon orderly se-

quence, such as learning all "dab" letters
and all "dit" letters in separate groups, is to
be discouraged. Before beginning with a code
practice set it is necessary to memorize the
whole alphabet perfectly. A good plan is to
study only two or three letters a day and to
drill with those letters until they become part
of your consciousness. Mentally translate each
day's letters into their sound equivalent
wherever they are seen, on signs, in papers,
indoors and outdoors. Tackle two additional
letters in the code chart each day, at the same
time reviewing the characters already learned.
Avoid memorizing by routine. Be able to
sound out any letter immediately without so
much as hesitating to think about the letters
preceding or following the one in question.
Know C, for example, apart from the sequence
ABC. Skip about among all the characters
learned, and before very long sufficient letters
will have been acquired to enable you to spell
out simple words to yourself in "dit dabs."
This is interesting exercise, and for that reason it is good to memorize all the vowels first
and the most common consonants next.
Actual code practice should start only when
the entire alphabet, the numerals, period, corn-

www.americanradiohistory.com

Introduction to Radio

16

THE

RADIO

tion, do it in code. It makes more interesting
practice than confining yourself to random
practice material.
hen two co- learners have memorized the
code and are ready to start sending to each
other for practice, it is a good idea to enlist
the aid of an experienced operator for the first
practice session or two so that they will get
an idea of how properly formed characters
sound.
Figure 2
These code characters are used in languages
other than English. They may occasionally
be encountered so it is well to know them.

ma, and question mark have been memorized
so thoroughly that any one can be sounded
without the slightest hesitation. Do not bother

with other punctuation or miscellaneous signals until later.

-

Each letter and figure must be
memorized by its sound rather
than its appearance. Code is a
system of sound communication, the same as
is the spoken word. The letter A, for example,
is one short and one long sound in combination sounding like dit dab, and it must be remembered as such, and not as "dot dash."
Sound
Not Sight

Practice

Time, patience, and regularity are
required to learn the code properly.
Do not expect to accomplish it within a few
days.
Don't practice too long at one stretch; it
does more harm than good. Thirty minutes at
a time should be the limit.
Lack of regularity in practice is the most
common cause of lack of progress. Irregular
practice is very little better than no practice
at

all. Write down what you have heard; then

forget it; do not look back. If your mind dwells
even for an instant on a signal about which
you have doubt, you will miss the next few
characters while your attention is diverted.
While various automatic code machines,
phonograph records, etc., will give you practice, by far the best practice is to obtain a
study companion who is also interested in
learning the code. When you have both memorized the alphabet you can start sending to
each other. Practice with a key and oscillator
or key and buzzer generally proves superior
to all automatic equipment. Two such sets
operated between two rooms are fine -or between your house and his will be just that
much better. Avoid talking to your partner
while practicing. If you must ask him a ques-

During the first practice period the speed
should be such that substantially solid copy
can be made without strain. Never mind if this
is only two or three words per minute. In the
next period the speed should be increased
slightly to a point where nearly all of the
characters can be caught only through conscious effort. When the student becomes proficient at this new speed, another slight increase may be made, progressing in this manner until a speed of about 16 words per minute
is attained if the object is to pass the amateur
13 -word per minute code test. The margin of
3 w.p.m. is recommended to overcome a possible excitement factor at examination time.
Then when you take the test you don't have to
worry about the "jitters" or an "off day."
Speed should not be increased to a new
level until the student finally makes solid
copy with ease for at least a five -minute
period at the old level. How frequently increases of speed can be made depends upon
individual ability and the amount of practice.
Each increase is apt to prove disconcerting,
but remember "you are never learning when

you are comfortable."
A number of amateurs are sending code
practice on the air on schedule once or twice
each week; excellent practice can be obtained
after you have bought or constructed your re-

ceiver by taking advantage of these sessions.
If you live in a medium -size or large city,
the chances are that there is an amateur radio
club in your vicinity which offers free code
practice lessons periodically.
Skill

listen to someone speaking
you do not consciously think how his
words are spelled. This is also true when you
read. In code you must train your ears to read
code just as your eyes were trained in school
to read printed matter. With enough practice
you acquire skill, and from skill, speed. In
other words, it becomes a habit, something
which can be done without conscious effort.
Conscious effort is fatal to speed; we can't
think rapidly enough; a speed of 25 words a
minute, which is a common one in commercial
operations, means 125 characters per minute
or more than two per second, which leaves
no time for conscious thinking.
When you

www.americanradiohistory.com

Learning the Code

HANDBOOK
Perfect Formation
of Characters

When

transmitting on the

code practice set to your
partner, concentrate on the
quality of your sending, not on your speed.
Your partner will appreciate it and he could
not copy you if you speeded up anyhow.
If you want to get a reputation as having an
excellent "fist" on the air, just remember that
speed alone won't do the trick. Proper execution of your letters and spacing will make
much more of an impression. Fortunately, as
you get so that you can send evenly and accurately, your sending speed will automatically
increase. Remember to try to see how evenly
you can send, and how fast you can receive.
Concentrate on making signals properly with
your key. Perfect formation of characters is
paramount to everything else. Make every signal right no matter if you have to practice it
hundreds or thousands of times. Never allow
yourself to vary the slightest from perfect formation once you have learned it.
If possible, get a good operator to listen to
your sending for a short time, asking him to
criticize even the slightest imperfections.
Timing
It is of the utmost importance to
maintain uniform spacing in characters and combinations of characters. Lack of
uniformity at this point probably causes beginners more trouble than any other single factor. Every dot, every dash, and every space
must be correctly timed. In other words, accurate timing is absolutely essential to intelligibility, and timing of the spaces between
the dots and dashes is just as important as
the lengths of the dots and dashes themselves.
The characters are timed with the dot as a
"yardstick." A standard dash is three times
as long as a dot. The spacing between parts
of the same letter is equal to one dot; the
space between letters is equal to three dots,
and that between words equal to five dots.
The rule for spacing between letters and
words is not strictly observed when sending
slower than about 10 words per minute for the
benefit of someone learning the code and desiring receiving practice. When sending at,
say, 5 w.p.m., the individual letters should be
made the same as if the sending rate were
about 10 w.p.m., except that the spacing between letters and words is greatly exaggerated.
The reason for this is obvious. The letter L,
for instance, will then sound exactly the same
at 10 w.p.m. as at 5 w.p.m., and when the
speed is increased above 5 w.p.m. the student
will not have to become familiar with what
may seem to him like a new sound, although
it is in reality only a faster combination of
dots and dashes. At the greater speed he will
merely have to learn the identification of the
same sound without taking as long to do so.

17

Or-:C,t,.

0oó000boá

taMS

ins

C

B

tmo

tit tt> riti

A

O
Figure

IMP

N

E

3

Diagram illustrating relative lengths of
dashes and spaces referred to the duration
of o dot. A dash is exactly equal in duration
to three dots; spaces between parts of a
letter equal one dot; those between letters,
three dots; space between words, five dots.
Note that a slight increase between two parts
of a letter will make it sound like two

letters.

Be particularly careful of

letters like

B.

Many beginners seem to have a tendency to

leave a longer space after the dash than that
which they place between succeeding dots,
thus making it sound like TS. Similarly, make
sure that you do not leave a longer space after
the first dot in the letter C than you do between other parts of the same letter; otherwise
it will sound like NN.
Once you have memorized the
code thoroughly you should concentrate on increasing your receiving speed. True, if you have to practice
with another newcomer who is learning the
code with you, you will both have to do some
sending. But don't attempt to practice sending
just for the sake of increasing your sending
speed.
When transmitting on the code practice set
to your partner so that he can get receiving
practice, concentrate on the quality of your
sending, not on your speed.
Because it is comparatively easy to learn
to send rapidly, especially when no particular
care is given to the quality of sending, many
operators who have just received their licenses
get on the air and send mediocre or worse code
at 20 w.p.m. when they can barely receive
good code at 13. Most oldtimers remember their
own period of initiation and are only too glad
to be patient and considerate if you tell them
that you are a newcomer. But the surest way
to incur their scorn is to try to impress them
with your "lightning speed," and then to request them to send more slowly when they
come back at you at the same speed.
Stress your copying ability; never stress
your sending ability. It should be obvious that
that if you try to send faster than you can receive, your ear will not recognize any mistakes which your hand may make.
Sending vs.

Receiving

www.americanradiohistory.com

1

8

I

n

t ro d u c t

i

o n

t

o

R a d

i

T

o

H E

R A D

I

O

fingers to become tense. Send with a full, free
arm movement. Avoid like the plague any finger motion other than the slight cushioning
effect mentioned above.
Stick to the regular hand key for learning
code. No other key is satisfactory for this purpose. Not until you have thoroughly mastered
both sending and receiving at the maximum
speed in which you are interested should you
tackle any form of automatic or semi -automatic
key such as the Vibroplex ( "bug ") or an electronic key.
Difficulties
Figure 4
PROPER POSITION OF THE FINGERS FOR
OPERATING A TELEGRAPH KEY
The fingers hold the knob and act os a cushion. The hand rests lightly on the key. The
muscles of the forearm provide the power,
the wrist acting as the fulcrum. The power
should not come from the fingers, but rather
from the forearm muscles.

Figure 4 shows the proper position of the hand, fingers and
wrist when manipulating a telegraph or radio
key. The forearm should rest naturally on the
desk. It is preferable that the key be placed
far enough back from the edge of the table
(about 18 inches) that the elbow can rest on
the table. Otherwise, pressure of the table
edge on the arm will tend to hinder the circulation of the blood and weaken the ulnar nerve
at a point where it is close to the surface,
which in turn will tend to increase fatigue
considerably.
The knob of the key is grasped lightly with
the thumb along the edge; the index and third
fingers rest on the top towards the front or far
edge. The hand moves with a free up and down
motion, the wrist acting as a fulcrum. The
power must come entirely from the arm muscles. The third and index fingers will bend
slightly during the sending but not because of
deliberate effort to manipulate the finger muscles. Keep your finger muscles just tight
enough to act as a cushion for the arm motion
and let the slight movement of the fingers take
care of itself. The key's spring is adjusted to
the individual wrist and should be neither too
stiff nor too loose. Use a moderately stiff tension at first and gradually lighten it as you
become more proficient. The separation between the contacts must be the proper amount
for the desired speed, being somewhat under
1/16 inch for slow speeds and slightly closer
together (about 1/32 inch) for faster speeds.
Avoid extremes in either direction.
Do not allow the muscles of arm, wrist, or
Using the Key

Should you experience difficulty
in increasing your code speed
after you have once memorized the characters,
there is no reason to become discouraged. It
is more difficult for some people to learn code
than for others, but there is no justification
for the contention sometimes made that "some
people just can't learn the code." It is not a
matter of intelligence; so don't feel ashamed
if you seem to experience a little more than
the usual difficulty in learning code. Your reaction time may be a little slower or your coordination not so good. If this is the case,
remember you can still learn the code. You
may never learn to send and receive at 40
w.p.m., but you can learn sufficient speed for
all non -commercial purposes and even for most
commercial purposes if you have patience,
and refuse to be discouraged by the fact that
others seem to pick it up more rapidly.
When the sending operator is sending just
a bit too fast for you (the best speed for practice), you will occasionally miss a signal or a
small group of them. When you do, leave a
blank space; do not spend time futilely trying
to recall it; dismiss it, and center attention
on the next letter; otherwise you'll miss more.
Do not ask the sender any questions until the

transmission is finished.
To prevent guessing and get equal practice
on the less common letters, depart occasionally from plain language material and use a jumble of letters in which the usually less commonly used letters predominate.
As mentioned before, many students put a
greater space after the dash in the letter B
than between other parts of the same letter so

it

sounds like TS.

C,

F, Q,

V,

X, Y and Z

often give similar trouble. Make a list of words
or arbitrary combinations in which these letters predominate and practice them, both sending and receiving until they no longer give you
trouble. Stop everything e l s e and stick at
them. So long as they give you trouble you are
not ready for anything else.
Follow the same procedure with letters
which you may tend to confuse such as F and
L, which are often confused by beginners.

www.americanradiohistory.com

HANDBOOK
Figure

Learning the Code

19

5

THE SIMPLEST CODE PRACTICE
SET CONSISTS OF A KEY AND A
INEXPENSIVE 500
OHM POTENTIOMETER
VOLUME CONTROL

BUZZER
is adjusted to give a
steady, high -pitched whine. If desired, the phones may be omitted,
in which case the buzzer should be
mounted firmly on a sounding board.
Crystal, magnetic, or dynamic earphones may be used. Additional
sets of phones should be connected
in parallel, not in series.
The buzzer

=

1.5 TO
S VOLTS
OF BATTERY

1

KEY

Keep at it until you always get them right
without having to stop even an instant to think
about it.
If you do not instantly recognize the sound
of any character, you have not learned it; go
back and practice your alphabet further. You
should never have to omit writing down every
signal you hear except when the transmission
is too fast for you.
Write down what you hear, not what you
think it should be. It is surprising how often
the word which you guess will be wrong.

All good operators copy several words behind, that is,
while one word is being received, they are
writing down or typing, say, the fourth or fifth
previous word. At first this is very difficult,
but after sufficient practice it will be found
actually to be easier than copying close up.
It also results in more accurate copy and enables the receiving operator to capitalize and
Copying Behind

CH-722

COLLECTOR
2= BASE
3= EMITTER
REO

00T

2000 n
PHONES

10K

KEY

0.5 W

Figure

PHONES.
TO 4
PAIR

6

SIMPLE TRANSISTOR CODE
PRACTICE OSCILLATOR
An inexpensive Raytheon CK -722 transistor
requires only a single 11,2 -volt flashlight
battery for power. The inductance of the earphone windings forms part of the oscillatory
circuit. The pitch of the note may be changed
by varying the value of the two capacitors
connected across the earphones.

THESE PARTS REQUIRED
ONLY IF HEADPHONE
OPERATION IS DESIRED

punctuate copy as he goes along. It is not recommended that the beginner attempt to do this
until he can send and receive accurately and
with ease at a speed of at least 12 words a
minute.
It requires a considerable amount of training to dissociate the action of the subconscious mind from the direction of the conscious
mind. It may help some in obtaining this training to write down two columns of short words.
Spell the first word in the first column out loud
while writing down the first word in the second
column. At first this will be a bit awkward,
but you will rapidly gain facility with practice.
Do the same with all the words, and then re-

verse columns.
Next try speaking aloud the words in the one
column while writing those in the other column;
then reverse columns.
After the foregoing can be done easily, try
sending with your key the words in one column while spelling those in the other. It won't
be easy at first, but it is well worth keeping
after if you intend to develop any real code
proficiency. Do not attempt to catch up. There
is a natural tendency to close up the gap, and
you must train yourself to overcome this.
Next have your code companion send you a
word either from a list or from straight text;
do not write it down yet. Now have him send
the next word; after receiving this second
word, write down the first word. After receiving the third word, write the second word; and
so on. Never mind how slowly you must go,
even if it is only two or three words per minute.
Stay behind.
It will probably take quite a number of practice sessions before you can do this with any
facility. After it is relatively easy, then try
staying two words behind; keep this up until
it is easy. Then try three words, four words,
and five words. The more you practice keeping received material in mind, the easier it
will be to stay behind. It will be found easier
at first to copy material with which one is
fairly familiar, then gradually switch to less
familiar material.

www.americanradiohistory.com

20

Introduction

to

R

adio

The two practice sets which
are described in this chapter
are of most value when you
have someone with whom to practice. Automatic code machines are not recommended to anyone who can possibly obtain a companion with
whom to practice, someone who is also interested in learning the code. If you are unable
to enlist a code partner and have to practice
Automatic Code
Machines

by yourself, the best way to g e t receiving
practice is by the use of a tape machine (automatic code sending machine) with several
practice tapes. Or you can use a set of phonograph code practice records. The records are
of use only if you have a phonograph whose
turntable speed is readily adjustable. The tape
machine can be rented by the month for a reasonable fee.
Once you can copy about 10 w.p.m. you can
also get receiving practice by listening to slow
sending stations on your receiver. Many amateur stations send slowly particularly when
working far distant stations. When receiving
conditions are particularly poor many commercial stations also send slowly, sometimes repeating every word. Until you can copy around
10 w.p.m. your receiver isn't much use, and
either another operator or a machine or records
are necessary for getting receiving practice
after you have once memorized the code.
Code Practice
Sets

If you don't feel too foolish
doing it, you can secure a
measure of code practice with

the help of a partner by sending "dit-dah"
messages to each other while riding to work,
eating lunch, etc. It is better, however, to use
a buzzer or code practice oscillator in conjunction with a regular telegraph key.
As a good key may be considered an investment it is wise to make a well -made key your
first purchase. Regardless of what type code
practice set you use, you will need a key, and
later on you will need one to key your trans-

mitter. If you get a good key to begin with,
you won't have to buy another one later.
The key should be rugged and have fairly
heavy contacts. Not only will the key stand
up better, but such a key will contribute to
the "heavy" type of sending so desirable for
radio work. Morse (telegraph) operators use
a "light" style of sending and can send somewhat faster when using this light touch. But,
in radio work static and interference are often
present, and a slightly heavier dot is desirable. If you use a husky key, you will find
yourself automatically sending in this manner.
To generate a tone simulating a code signal
as heard on a receiver, either a mechanical
buzzer or an audio oscillator may be used. Figure 5 shows a simple code-practice set using
a buzzer which may be used directly simply
by mounting the buzzer on a sounding board,
or the buzzer may be used to feed from one to
four pairs of conventional high -impedance
phones.
An example of the audio -oscillator type of
code -practice set is illustrated in figures 6
and 7. An inexpensive Raytheon CK -722 transistor is used in place of the more expensive,
power consuming vacuum tube. A single "pen lite" 1i-volt cell powers the unit. The coils
of the earphones form the inductive portion
of the resonant circuit. 'Phones having an
impedance of 2000 ohms or higher should be
used. Surplus type R -14 earphones also work
well with this circuit.

Figure

7

circuit of Figure 6 is used in this
miniature transistorized code Practice
oscillator. Components are mounted in a
small plastic case. The transistor is
The

attached to a three terminal phenolic
mounting strip. Sub- miniature jacks are
used for the key and phones connections.
A hearing aid earphone may also be used,
as shown. The phone is stored in the

plastic case when not in use.

www.americanradiohistory.com

CHAPTER TWO

Direct Current Circuits

so different particles, but this further subdivision can be left to quantum mechanics and
atomic physics. As far as the study of electronics is concerned it is only necessary for
the reader to think of the normal atom as being
composed of a nucleus having a net positive
charge that is exactly neutralized by the one
or more orbital electrons surrounding it.
The atoms of different elements differ in
respect to the charge on the positive nucleus
and in the number of electrons revolving
around this charge. They range all the way
from hydrogen, having a net charge of one
on the nucleus and one orbital electron, to
uranium with a net charge of 92 on the nucleus
and 92 orbital electrons. The number of orbital
electrons is called the atomic number of the
element.

All naturally occurring matter (excluding
artifically produced radioactive substances) is
made up of 92 fundamental constituents called
elements. These elements can exist either in
the free state such as iron, oxygen, carbon,
copper, tungsten, and aluminum, or in chemical unions commonly called compounds. The
smallest unit which still retains all the original characteristics of an element is the atom.
Combinations of atoms, or subdivisions of
compounds, result in another fundamental
unit, the molecule. The molecule is the smallest unit of any compound. All reactive elements when in the gaseous state also exist
in the molecular form, made up of two or more

atoms. The nonreactive gaseous elements
helium, neon, argon, krypton, xenon, and
radon are the only gaseous elements that ever
exist in a stable monatomic state at ordinary
temperatures.

From the above it must not be
thought that the electrons revolve in a haphazard manner
around the nucleus. Rather, the electrons in
an element having a large atomic number are
grouped into rings having a definite number of
electrons. The only atoms in which these rings
Action of the
Electrons

The Atom

2-1

An atom is an extremely small unit of
matter there are literally billions of them
making up so small a piece of material as a
speck of dust. To understand the basic theory
of electricity and hence of radio, we must go
further and divide the atom into its main
components, a positively charged nucleus and
a cloud of negatively charged particles that
surround the nucleus. These particles, swirling
around the nucleus in elliptical orbits at an
incredible rate of speed, are called orbital

-

are completely filled are those of the inert
gases mentioned before; all other elements
have one or more uncompleted rings of electrons. If the uncompleted ring is nearly empty,
the element is metallic in character, being
most metallic when there is only one electron
in the outer ring. If the incomplete ring lacks
only one or two electrons, the element is
usually non- metallic. Elements with a ring
about half completed will exhibit both nonmetallic and metallic characteristics; carbon,
silicon, germanium, and arsenic are examples.
Such elements are called semi- conductors.
In metallic elements these outer ring electrons are rather loosely held. Consequently,

electrons.
It is upon the behavior of these electrons
when freed from the atom, that depends the
study of electricity and radio, as well as
allied sciences. Actually it is possible to subdivide the nucleus of the atom into a dozen or

21

www.americanradiohistory.com

22

Direct Current Circuits

there is a continuous helter -skelter movement
of these electrons and a continual shifting
from one atom to another. The electrons which
move about in a substance are called free
electrons, and it is the ability of these electrons to drift from atom to atom which makes
possible the electric current.

If

the free electrons are numerous and loosely held, the
element is a good conductor.
On the other hand, if there are few free electrons, as is the case when the electrons in an
outer ring are tightly held, the element is a
poor conductor. If there are virtually no free
electrons, the element is a good insulator.
Conductors and
Insulators

2-2

Fundamental Electrical

Units and Relationships
Electromotive Force:
Potential Difference

The free electrons in
a conductor move constantly about and change
their position in a haphazard manner. To
produce a drift of electrons or electric current
along a wire it is necessary that there be a
difference in "pressure" or potential between
the two ends of the wire. This potential difference can be produced by connecting a
source of electrical potential to the ends of
the wire.
As will be explained later, there is an excess of electrons at the negative terminal of
a battery and a deficiency of electrons at the
positive terminal, due to chemical action.
When the battery is connected to the wire, the
deficient atoms at the positive terminal attract
free electrons from the wire in order for the
positive terminal to become neutral. The
attracting of electrons continues through the
wire, and finally the excess electrons at
the negative terminal of the battery are attracted by the positively charged atoms at the
end of the wire. Other sources of electrical
potential (in addition to a battery) are: an
electrical generator (dynamo), a thermocouple,
an electrostatic generator (static machine), a
photoelectric cell, and a crystal or piezoelectric generator.
Thus it is seen that a potential difference
is the result of a difference in the number of
electrons between the two (or more) points in
question. The force or pressure due to a
potential difference is termed the electromotive force, usually abbreviated e. m. f. or
E.M.F. It is expressed in units called volts.
It should be noted that for there to be a
potential difference between two bodies or
points it is not necessary that one have a
positive charge and the other a negative
charge. If two bodies each have a negative

THE

RADIO

charge, but one more negative than the other,
the one with the lesser negative charge will
act as though it were positively charged with
respect to the other body. It is the algebraic
potential difference that determines the force
with which electrons are attracted or repulsed,
the potential of the earth being taken as the
zero reference point.

The flow of electrons along a
conductor due to the application
of an electromotive force constitutes an electric current. This drift is in
addition to the irregular movements of the
electrons. However, it must not be thought
that each free electron travels from one end
of the circuit to the other. On the contrary,
each free electron travels only a short distance
before colliding with an atom; this collision
generally knocking off one or more electrons
from the atom, which in turn move a short
distance and collide with other atoms, knocking off other electrons. Thus, in the general
drift of electrons along a wire carrying an
electric current, each electron travels only a
short distance and the excess of electrons at
one end and the deficiency at the other are
balanced by the source of the e.m.f. When this
source is removed the state of normalcy returns; there is still the rapid interchange of
free electrons between atoms, but there is no
general trend or "net movement" in either
one direction or the other.
The Electric
Current

There are two units of measure ment associated with current,
and they are often confused.
The rate of flou of electricity is stated in
amperes. The unit of quantity is the coulomb.
A coulomb is equal to 6.28 x 10" electrons,
and when this quantity of electrons flows by
a given point in every second, a current of
one ampere is said to be flowing. An ampere
is equal to one coulomb per second; a coulomb
is, conversely, equal to one ampere- second.
Thus we see that coulomb indicates amount,
and ampere indicates rate of flow of electric
current.
Older textbooks speak of current flow as
being from the positive terminal of the e.m.f.
source through the conductor to the negative
terminal. Nevertheless, it has long been an
established fact that the current flow in a
metallic conductor is the electronic flow from
the negative terminal of the source of voltage
through the conductor to the positive terminal.
The only exceptions to the electronic direction
of flow occur in gaseous and electrolytic conductors where the flow of positive ions toward
the cathode or negative electrode constitutes
a positive flow in the opposite direction to the
electronic flow. (An ion is an atom, molecule,
Ampere and
Coulomb

www.americanradiohistory.com

HANDBOOK

Resistance

or particle which either lacks one or more
electrons, or else has an excess of one or
more electrons.)
In radio work the terms "electron flow" and

"current" are becoming accepted as being
synonymous, but the older terminology is still
accepted in the electrical (industrial) field.
Because of the confusion this sometimes
causes, it is often safer to refer to the direction of electron flow rather than to the direction of the "current." Since electron flow
consists actually of a passage of negative
charges, current flow and algebraic electron
flow do pass in the same direction.
The flow of current in a material
depends upon the ease with
which electrons can be detached from the
atoms of the material and upon its molecular
structure. In other words, the easier it is to
detach electrons from the atoms the more free
electrons there will be to contribute to the
flow of current, and the fewer collisions that
occur between free electrons and atoms the
greater will be the total electron flow.
The opposition to a steady electron flow
is called the resistance of a material, and is
one of its physical properties.
The unit of resistance is the ohm. Every

23

TABLE OF RESISTIVITY
'

Material
Aluminum
Bross

Cadmium
Chromium
Copper
Iron
Silver
Zinc
Nichrome
Const

Manganin
Monet

esist vny in
Ohms per

Temp. Coeff. of
resistance per =C
at 20° C.

Circular
Mil -Foot

0.0049
0.003 to 0.007
0.0038
0.00
0.0039
0.006
0.004
0.0035
0.0002

17

45
46
16

10.4
59
9.8
36

650

0.00001
0.00001
0.0019

295
290
255

FIGURE

1

Resistance

substance has a specific resistance, usually
expressed as ohms per mil -foot, which is determined by the material' s molecular structure
and temperature. A mil -foot is a piece of
material one circular mil in area and one foot
long. Another measure of resistivity frequently
used is expressed in the units microhms per
centimeter cube. The resistance of a uniform
length of a given substance is directly proportional to its length and specific resistance,
and inversely proportional to its cross- sectional area. A wire with a certain resistance for a
given length will have twice as much resistance if the length of the wire is doubled. For
a given length, doubling the cross -sectional
area of the wire will halve the resistance,
while doubling the diameter will reduce the
resistance to one fourth. This is true since
the cross -sectional area of a wire varies as
the square of the diameter. The relationship
between the resistance and the linear dimensions of a conductor may be expressed by the
following equation:
R

R =
r =
l=
A =

Conductors and

resistance in ohms
resistivity in Ohms per mil-foot
length of conductor in feet
cross - sectional area in circular mils

In the

molecular structure of

glass,
porcelain, and mica all electrons are tightly held within their orbits and
there are comparatively few free electrons.
This type of substance will conduct an electric current only with great difficulty and is
known as an insulator. An insulator is said to
have a high electrical resistance.
On the other hand, materials that have a
large number of free electrons are known as
conductors. Alost metals, those elements which
have only one or two electrons in their outer
ring, are good conductors. Silver, copper, and
aluminum, in that order, are the best of the
common metals used as conductors and are
said to have the greatest conductivity, or lowest resistance to the flow of an electric
current.
These units are the volt,
Fundamental

Insulators

many materials such as

the ampere, and the ohm.
They were mentioned in the
preceding paragraphs, but were not completely
defined in terms of fixed, known quantities.
The fundamental unit of current, or rate of
flow of electricity is the ampere. A current of
one ampere will deposit silver from a specified solution of silver nitrate at a rate of
1.118 milligrams per second.
Electrical Units

=-rl
A

Where

The resistance also depends upon temperature, increasing with increases in temperature
for most substances (including most metals),
due to increased electron acceleration and
hence a greater number of impacts between
electrons and atoms. However, in the case of
some substances such as carbon and glass the
temperature coefficient is negative and the
resistance decreases as the temperature increases. This is also true of electrolytes. The
temperature may be raised by the external application of heat, or by the flow of the current
itself. In the latter case, the temperature is
raised by the heat generated when the electrons
and atoms collide.

www.americanradiohistory.com

24

Direct Current Circuits

THE

RADIO

1111111111111111

lu
1

Figure

2

TYPICAL RESISTORS
Shown above are various types of resistors used in electronic circuits. The larger units are
power resistors. On the left is a variable power resistor. Three precision -type resistors ore
shown in the tenter with two small composition resistors beneath them. At the right is o

composition -type potentiometer, used for audio circuitry.

The international standard for the ohm is
the resistance offered by a uniform column of
mercury at 0°C., 14.4521 grams in mass, of
constant cross - sectional area and 106.300
centimeters in length. The expression megohm
(1,000,000 ohms) is also sometimes used
when speaking of very large values of resistance.
A volt is the e.m.f. that will produce a current of one ampere through a resistance of
one ohm. The standard of electromotive force
is the Weston cell which at 20 °C. has a
potential of 1.0183 volts across its terminals.
This cell is used only for reference purposes
in a bridge circuit, since only an infinitesimal

-vw-v-

RESISTANCE

Ri

vor

CONDUCTORS

B2

_-

BATTERY

E

Figure

3

SIMPLE SERIES CIRCUITS
At (A) the battery is in series with a single
resistor. At (B) the battery is in series with
two resistors, the resistors themselves being
in series. The arrows indicate the direction of
electron flow.

amount of current may be drawn from it without disturbing its characteristics.

The relationship between the
electromotive force (voltage),
the flow of current (amperes), and the resistance which impedes the flow of current (ohms),
is very clearly expressed in a simple but
highly valuable law known as Ohm's laun.
This law states that the current in amperes is
equal to the voltage in volts divided by the
resistance in ohms. Expressed as an equation:
Ohm's Law

I

=-RE

If the voltage (E) and resistance (R) are
known, the current (I) can be readily found.
If the voltage and current are known, and the
resistance is unknown, the resistance (R) is

E

equal to

.

When the

voltage is the un-

known quantity, it can be found by multiplying I x R. These three equations are all secured
from the original by simple transposition.
The expressions are here repeated for quick

reference:

E

I

=R

www.americanradiohistory.com

R=-E
I

E = IR

Resistive Circuits

HANDBOOK

Figure 4
SIMPLE PARALLEL

CIRCUIT

The two resistors RI and R2 are said to be in
parallel since the flow of current is offered
two parallel paths. An electron leaving point
A will pass either through R1 or R2, but not
through both, to reach the positive terminal
of the battery. If a large number of lectrons
are considered, the greater number will pass
through whichever of the two resistors has
the lower resistance.

where I is the current in amperes,
R is the resistance in ohms,
E is the electromotive force in volts.

Application of

All electrical circuits fall in-

Ohm's Law

to one of three classes: series
circuits, parallel circuits, and

series -parallel circuits. A series circuit is
one in which the current flows in a single
continuous path and is of the same value at
every point in the circuit (figure 3). In a parallel circuit there are two or more current
paths between two points in the circuit, as
shown in figure 4. Here the current divides at
A, part going through R, and part through R2i
and combines at B to return to the battery.
Figure 5 shows a series -parallel circuit. There
are two paths between points A and B as in
the parallel circuit, and in addition there are
two resistances in series in each branch of
the parallel combination. Two other examples
of series -parallel arrangements appear in figure 6. The way in which the current splits to
flow through the parallel branches is shown by
the arrows.
In every circuit, each of the parts has some
resistance: the batteries or generator, the connecting conductors, and the apparatus itself.
Thus, if each part has some resistance, no
matter how little, and a current is flowing
through it, there will be a voltage drop across
it. In other words, there will be a potential
difference between the two ends of the circuit
element in question. This drop in voltage is
equal to the product of the current and the
resistance, hence it is called the IR drop.
The source of voltage has an internal resistance, and when connected into a circuit
so that current flows, there will be an IR drop
in the source just as in every other part of the
circuit. Thus, if the terminal voltage of the
source could be measured in a way that would
cause no current to flow, it would be found
to be more than the voltage measured when a
current flows by the amount of the IR drop

25

Figure 5
SERIES-PARALLEL
CIRCUIT

In this type of circuit the resistors are arranged in series groups, and these serlesed
groups ore then placed in parallel.

in the source. The voltage measured with no
current flowing is termed the no load voltage;
that measured with current flowing is the load
voltage. It is apparent that a voltage source
having a low internal resistance is most de-

sirable.
The current flowing in a series
circuit is equal to the voltage
impressed divided by the total
resistance across which the voltage is impressed. Since the same current flows through
every part of the circuit, it is merely necessary to add all the individual resistances to
obtain the total resistance. Expressed as a
formula:
Resistances
in Series

+...

+RN .
Riotai =RI +R2 +R,
if the resistances happened to be
all the same value, the total resistance would
be the resistance of one multiplied by the
number of resistors in the circuit.
Of course,

Consider two resistors, one of
100 ohms and one of 10 ohms,
connected in parallel as in figure 4, with a voltage of 10 volts applied
across each resistor, so the current through
each can be easily calculated.
Resistances
in Parallel

E

I= -R
E = 10

volts

I, =

R = 100 ohms
E = 10

volts

R

ohms

10

Total current

10

= 0.1 ampere
100

-= 1.0 ampere
10

I2

=

10

=

I, +

12

= 1.1 ampere

Until it divides at A, the entire current of
1.1 amperes is flowing through the conductor
from the battery to A, and again from B through
the conductor to the battery. Since this is more
current than flows through the smaller resistor
it is evident that the resistance of the parallel
combination must be less than 10 ohms, the
resistance of the smaller resistor. We can find
this value by applying Ohm's law.

www.americanradiohistory.com

Direct Current Circuits

26

THE

R=-E

RADIO

A

I

E = 10
I

= 1.1

volts
amperes

10
R

=

1.1

9.09 ohms

The resistance of the parallel combination is
9.09 ohms.
Mathematically, we can derive a simple
formula for finding the effective resistance of
two resistors connected in parallel.
This formula is:

-

R

where

R

R,
R2

R, x R,

+R,
is the unknown resistance,
is the resistance of the first resistor,
is the resistance of the second resistor.

If the effective value required is known,
and it is desired to connect one unknown resistor in parallel with one of known value,
the following transposition of the above formula will simplify the problem of obtaining
the unknown value:

R2

where

R, x R

-

R,

-R

is the effective value required,
R, is the known resistor,
R2 is the value of the unknown resistance necessary to give R when
in parallel with R,.
R

The resultant value of placing a number of
unlike resistors in parallel is equal to the reciprocal of the sum of the reciprocals of the
various resistors. This can be expressed as:

R=

1

- +1

R,

1

R,

Figure 6
OTHER COMMON SERIES -PARALLEL
CIRCUITS

R,

-+...
R,
1

+

-

resistors connected in parallel is always
less than the value of the lowest resistance in
more

the combination. It is well to bear this simple
rule in mind, as it will assist greatly in approximating the value of paralleled resistors.
To find the total resistance of
several resistors connected in
series -parallel, it is usually
easiest to apply either the formula for series
resistors or the parallel resistor formula first,
in order to reduce the original arrangement to
a simpler one. For instance, in figure 5 the
series resistors should be added in each
branch, then there will be but two resistors in
parallel to be calculated. Similarly in figure 7,
although here there will be three parallel resistors after adding the series resistors in
each branch. In figure 6B the paralleled resistors should be reduced to the equivalent
series value, and then the series resistance
values can be added.
Resistances in series -parallel can be solved
by combining the series and parallel formulas
into one similar to the following (refer to
figure 7):
Resistors in
Series Parallel

1

R.

R1

The effective value of placing any number
of unlike resistors in parallel can be determined from the above formula. However, it
is commonly used only when there are three
or more resistors under consideration, since
the simplified formula given before is more
convenient when only two resistors are being
used.
From the above, it also follows that when
two or more resistors of the same value are
placed in parallel, the effective resistance of
the paralleled resistors is equal to the value
of one of the resistors divided by the number
of resistors in parallel.
The effective value of resistance of two or

R,+

+--

1

1

R,

R, + R,

1

Rs

+R6+R,

A
voltage divider is exactly what its name implies: a resistor or a series of resistors connected across a source of voltage from which
various lesser values of voltage may be obtained by connection to various points along
the resistor.
A voltage divider serves a most useful purpose in a radio receiver, transmitter or amplifier, because it offers a simple means of
obtaining plate, screen, and bias voltages of
different values from a common power supply

Voltage Dividers

www.americanradiohistory.com

HANDBOOK

Voltage

Divider

27

BLEEDER CURRENT

i__

FLOWS BETWEEN
POINTS A AND B

EATERNAL
LOAD

Figure 7
ANOTHER TYPE OF
SERIES -PARALLEL CIRCUIT

Figure 8
SIMPLE VOLTAGE DIVIDER

source. It may also be used to obtain very low
voltages of the order of .01 to .001 volt with
a high degree of accuracy, even though a
means of measuring such voltages is lacking.
The procedure for making these measurements
can best be given in the following example.
Assume that an accurately calibrated voltmeter reading from 0 to 150 volts is available,
and that the source of voltage is exactly 100
volts. This 100 volts is then impressed through
a resistance of exactly 1,000 ohms. It will,
then, be found that the voltage along various
points on the resistor, with respect to the
grounded end, is exactly proportional to the
resistance at that point. From Ohm's law, the
current would be 0.1 ampere; this current remains unchanged since the original value of
resistance (1,000 ohms) and the voltage source
(100 volts) are unchanged. Thus, at a 500 ohm point on the resistor (half its entire resistance), the voltage will likewise be halved
or reduced to 50 volts.
The equation (E = I x R) gives the proof:
E = 500 x 0.1 = 50. At the point of 250 ohms
on the resistor, the voltage will be one -fourth
the total value, or 25 volts (E = 250 x 0.1 = 25).

Continuing with this process, a point can be
found where the resistance measures exactly
1 ohm and where the voltage equals 0.1 volt.
It is, therefore, obvious that if the original
source of voltage and the resistance can be
measured, it is a simple matter to predetermine the voltage at any point along the resistor, provided that the current remains constant,
and provided that no current is taken from the
tap -on point unless this current is taken into
consideration.
Proper design of a voltage
divider for any type of radio
equipment is a relatively
simple matter. The first consideration is the
amount of "bleeder current" to be drawn.
In addition, it is also necessary that the desired voltage and the exact current at each tap
on the voltage divider be known.
Figure 8 illustrates the flow of current in a
simple voltage divider and load circuit. The
light arrows indicate the flow of bleeder current, while the heavy arrows indicate the flow
of the load current. The design of a combined
Voltage Divider
Calculations

CIRCUIT
indicate the manner in which the
current flow divides between the voltage divider
itslf and th externo! load circuit.
The arrows

bleeder resistor and voltage divider, such as
is commonly used in radio equipment, is illustrated in the following example:
A power supply delivers 300 volts and is
conservatively rated to supply all needed current for the receiver and still allow a bleeder
current of 10 milliamperes. The following voltages are wanted: 75 volts at 2 milliamperes
for the detector tube, 100 volts at 5 milliamperes for the screens of the tubes, and
250 volts at 20 milliamperes for the plates of
the tubes. The required voltage drop across R,
is 75 volts, across R, 25 volts, across R, 150
volts, and across R, it is 50 volts. These
values are shown in the diagram of figure 9.
The respective current values are also indicated. Apply Ohm's law:
E

R,

75

=

= 7,500 ohms.

01

R,
R, =

R,

E

25

I

012

E
-_
I

-=
150

2,083 ohms.

8,823 ohms.

.017

50
=-E = .037

= 1,351 ohms.

RTotal = 7,500 + 2,083 + 8,823 +
1,351 = 19,757 ohms.
A 20,000 -ohm resistor with three sliding taps
will be of the approximately correct size, and
would ordinarily be used because of the diffi-

culty in securing four separate resistors of the
exact odd values indicated, and because no
adjustment would be possible to compensate
for any slight error in estimating the probable
currents through the various taps.
When the sliders on the resistor once are
set to the proper point, as in the above ex-

www.americanradiohistory.com

Direct Current Circuits

28

THE
-2

10 + 2 +5

+20

AMPS

1M

RI

MA.

50 VOLTS DROP

0V

0MA
A

-2-AMPS

L1,

R2

10 +2

+5 MA
150 VOLTS DROP

AMPi

-

1

300 VOLTS

+2

-1

(-/

10

RADIO

MA.

25 VOLTS DROP

Figure 10
ILLUSTRATING KIRCHHOFF'S

l

R CURRENT10 .A.J
BLEEDE75
VOLTS D,ROP

POWER SUPPLY

Figure

-

LOA

D

-

-

-

-

-

9

MORE COMPLEX VOLTAGE DIVIDER
The method for computing the values of the
resistors is discussed in the accompanying text.

FIRST LAW
The current flowing toward point "A" is
to the current flowing away from point

qual

"A."

-

sum of all currents flowing toward
and away from the point
taking signs into
account
is equal to zero. Such a sum is
known as an algebraic sum; such that the law
can be stated thus: The algebraic sum of all

tive, the

-

ample, the voltages will remain constant at
the values shown as long as the current remains a constant value.

currents entering and leaving a point is zero.
Figure 10 illustrates this first law. Since the
effective resistance of the network of resistors
is 5 ohms, it can be seen that 4 amperes flow

One of the serious disadvanrages of the voltage divider
becomes evident when the
the current drawn fromone of the taps changes.
It is obvious that the voltage drops are interdependent and, in turn, the individual drops
are in proportion to the current which flows
through the respective sections of the divider
resistor. The only remedy lies in providing a
heavy steady bleeder current in order to make
the individual currents so small a part of the
total current that any change in current will

toward point A, and 2 amperes flow away
through the two 5 -ohm resistors in series. The
remaining 2 amperes flow away through the 10ohm resistor. Thus, there are 4 amperes flowing
to point A and 4 amperes flowing away from
the point. If R is the effective resistance of
the network (5 ohms), R, = 10 ohms, R, = 5
ohms, R, = 5 ohms, and E = 20 volts, we can
set up the following equation:

Disadvantages of
Voltage Dividers

result in only

a

slightchange in voltage. This

can seldom be realized in practice because of
the excessive values of bleeder current which

would be required.

Kirchhoff's Laws

Ohm's law is all that is
necessary to calculate the
values in simple circuits, such as the preceding examples; but in more complex problems, involving several loops or more than
one voltage in the same closed circuit, the
use of Kirchhoff's laws will greatly simplify
the calculations. These laws are merely rules
for applying Ohm's law.
Kirchhoff's first law is concerned with net
current to a point in a circuit and states that:

At any point in a circuit the current
flowing toward the point is equal to
the current flowing away from the
point.
Stated in another way: if currents flowing to
the point are considered positive, and those
flowing from the point are considered nega-

E

E

R

R,

E

20

20

5

10

R2

+R,

=0

20
5

+5

4 -2 -2 =0
Kirchhoff's second law is concerned with

net voltage drop around a closed loop in

a

circuit and states that:
In any closed path or loop in a circuit
the sum of the IR drops must equal
the sum of the applied e.m. f.'s.

The second law also may be conveniently
stated in terms of an algebraic sum as: The
algebraic sum of all voltage drops around a
closed path or loop in a circuit is zero. The
applied e.m.f.'s (voltages) are considered
positive, while IR drops taken in the direction
of current flow (including the internal drop
of the sources of voltage) are considered
negative.
Figure 11 shows an example of the applica-

tion of Kirchhoff's laws to a comparatively
simple circuit consisting of three resistors and

www.americanradiohistory.com

Kirchoff's Laws

HANDBOOK

29

1 volt forces a current of 1 ampere
through a circuit. The power in a resistive
circuit is equal to the product of the voltage applied across, and the current flowing
in, a given circuit. Hence: P (watts) = E
(volts) x I (amperes).
Since it is often convenient to express
power in terms of the resistance of the circuit
and the current flowing through it, a substitution of IR for E (E = IR) in the above formula
gives: P = IR x I or P = 12R. In terms of voltage and resistance, P = E' /R. Here, I = E/R
and when this is substituted for I the original
formula becomes P = E x E /R, or P = E' /R.
To repeat these three expressions:

an e.m.f. of

1.

2.

SET VOLTAGE DROPS AROUND EACH LOOP EQUAL TO ZERO.

1121DHMS)+2(t

-12)+3 =0

-6+2 (12-11)

+312 °0

(FIRST LOOP)

(SECOND LOOP)

SIMPLIFY

211+211-212+3.0
411

+3

-

2

1

21a- 2It+31z -6 =0
512- 211 -6 =0

2

211+6
5

3.

411

+3

2

4

5

-

I

z

P =

EQUATE

2It +6

-

5

SIMPLIFY

and

2011+15= 411 +12
11 ß-t6 AMPERE

I

RE- SUBSTITUTE

Iz-

23

2

EI, P = I2R, and

P = E2 /R,

where P is the power in watts,
E is the electromotive force in volts,

t

2

I

á

AMPERE

Figure 11
ILLUSTRATING KIRCHHOFF'S
SECOND LAW
The voltage drop around any closed loop In a
network Is qual to zero.

two batteries. First assume an arbitrary direction of current flow in each closed loop of the
circuit, drawing an arrow to indicate the assumed direction of current flow. Then equate
the sum of all IR drops plus battery drops
around each loop to zero. You will need one
equation for each unknown to be determined.
Then solve the equations for the unknown currents in the general manner indicated in figure
11. If the answer comes out positive the direction of current flow you originally assumed
was correct. If the answer comes out negative,
the current flow is in the opposite direction to
the arrow which was drawn originally. This is
illustrated in the example of figure 11 where
the direction of flow of I, is opposite to the
direction assumed in the sketch.

In order to cause electrons
Resistive Circuits to flow through a conductor,
constituting a current flow,
it is necessary to apply an electromotive force
(voltage) across the circuit. Less power is
expended in creating a small current flow
through a given resistance than in creating
a large one; so it is necessary to have a unit
of power as a reference.
The unit of electrical power is the watt,
which is the rate of energy consumption when

is the current in amperes.

To apply the above equations to a typical
problem: The voltage drop across a cathode
resistor in a power amplifier stage is 50 volts;
the plate current flowing through the resistor
is 150 milliamperes. The number of watts the
resistor will be required to dissipate is found
from the formula: P = El, or 50 x .150 = 7.5
watts (.150 amperes is equal to 150 milliamperes). From the foregoing it is seen that
a 7.5 -watt resistor will safely carry the required current, yet a 10- or 20 -watt resistor
would ordinarily be used to provide a safety

factor.
In another problem, the conditions being
similar to those above, but with the resistance
(R = 333`/2 ohms), and current being the known
factors, the solution is obtained as follows:
P = I2R = .0225 x 333.33 = 7.5. If only the voltage and resistance are known, P = E2 /R =
2500/333.33 = 7.5 watts. It is seen that all
three equations give the same results; the
selection of the particular equation depends
only upon the known factors.

It

is important to remember
that power (expressed in watts,
horsepower, etc.), represents
the rate of energy consumption or the rate of
doing work. But when we pay our electric bill
Power, Energy
and Work

Power in

Figure 12
MATCHING OF
RESISTANCES

RL

I

To deliver the greatest amount of power to the
load, the load resistance RL should be equal to
the Internal reslstonce of the battery RI.

www.americanradiohistory.com

30

Direct Current Circuits

THE

RADIO

is said to have a certain capacitance. The
energy stored in an electrostatic field is expressed in joules (watt seconds) and is equal
to CE' /2, where C is the capacitance in farads
(a unit of capacitance to be discussed) and E
is the potential in volts. The charge is equal
to CE, the charge being expressed in coulombs.
metallic plates separated from each other by
a thin layer of insulating
material (called a dielectric, in this case),
becomes a capacitor. When a source of d-c
potential is momentarily applied across these
plates, they may be said to become charged.
If the same two plates are then joined together momentarily by means of a switch, the
capacitor will discharge.
When the potential was first applied, electrons immediately flowed from one plate to the
other through the battery or such source of
d -c potential as was applied to the capacitor
plates. However, the circuit from plate to
plate in the capacitor was incomplete (the two
plates being separated by an insulator) and
thus the electron flow ceased, meanwhile establishing a shortage of electrons on one plate
and a surplus of electrons on the other.
Remember that when a deficiency of electrons exists at one end of a conductor, there
is always a tendency for the electrons to move
about in such a manner as to re- establish a
state of balance. In the case of the capacitor
herein discussed, the surplus quantity of electrons on one of the capacitor plates cannot
move to the other plate because the circuit
has been broken; that is, the battery or d -c potential was removed. This leaves the capacitor in a charged condition; the capacitor plate
with the electron deficiency is positively
charged, the other plate being negative.
In this condition, a considerable stress
exists in the insulating material (dielectric)
which separates the two capacitor plates, due
to the mutual attraction of two unlike potentials on the plates. This stress is known as
electrostatic energy, as contrasted with electromagnetic energy in the case of an inductor.
This charge can also be called potential
energy because it is capable of performing
work when the charge is released through an
external circuit. The charge is proportional to
the voltage but the energy is proportional to
the voltage squared, as shown in the following
Capacitance and
Capacitors

sm.

nErg

Figure 13
TYPICAL CAPACITORS
large units ore high value filter capaci-

The two
tors. Shown beneath these ore various types of
by -pass capacitors for r-f and audio application.

to the power company we have purchased a
specific amount of energy or work expressed
in the common units of kilowatt- hours. Thus
rate of energy consumption (watts or kilowatts)
multiplied by time (seconds, minutes or hours)
gives us total energy or work. Other units of
energy are the watt- second, BTU, calorie, erg,
and joule.
Heating Effect

Heat is generated when a
source of voltage causes a
current to flow through a resistor (or, for that
matter, through any conductor). As explained
earlier, this is due to the fact that heat is
given off when free electrons collide with the
atoms of the material. More heat is generated
in high resistance materials than in those of
low resistance, since the free electrons must
strike the atoms harder to knock off other
electrons. As the heating effect is a function
of the current flowing and the resistance of
the circuit, the power expended in heat is
given by the second formula: P = I'R.
2 -3

Electrostatics

-

Capacitors

Electrical energy can be stored in an electrostatic field. A device capable of storing
energy in such a field is called capacitor
(in earlier usage the term condenser was
frequently used but the IRE standards call for
the use of capacitor instead of condenser) and

Two

analogy.
The charge represents a definite amount of
electricity, or a given number of electrons.
The potential energy possessed by these
electrons depends not only upon their number,
but also upon their potential or voltage.
Compare the electrons to water, and two

capacitors to standpipes, a

www.americanradiohistory.com

1

fifd.

capacitor to

Capacitance

HANDBOOK
ROSTATIC
A- EILECT
ELD

-

SHORTAGE
OF ELECTRONS

1

SURPLUS
OF ELECTRONS

1

Figure 14
SIMPLE CAPACITOR
Illustrating the imaginary lines of force repre
Renting the paths along which the repelling force

If the external circuit of
the two capacitor plates is
completed by joining the
terminals together with a piece of wire, the
electrons will rush immediately from one plate
to the other through the external circuit and
establish a state of equilibrium. This latter
phenomenon explains the discharge of a capacitor. The amount of stored energy in a charged
capacitor is dependent upon the charging potential, as well as a factor which takes into
account the size of the plates, dielectric
thickness, nature of the dielectric, and the
number of plates. This factor, which is determined by the foregoing, is called the capacitanceof a capacitor and is expressed in farads.
The farad is such a large unit of capacitance that it is rarely used in radio calculations, and the following more practical units
The Unit of Capacitance: The Farad

have, therefore, been chosen.
farad,

-

CxE'
2 x

1,000,000

This storage of energy in a capacitor is one
of its very important properties, particularly
in those capacitors which are used in power
supply filter circuits.

a standpipe having a cross section of 1 square
inch and a 2 pfd. capacitor to a standpipe having a cross section of 2 square inches. The
charge will represent a given volume of water,
as the "charge" simply indicates a certain
number of electrons. Suppose the water is
equal to 5 gallons.
Now the potential energy, or capacity for
doing work, of the 5 gallons of water will be
twice as great when confined to the 1 sq. in.
standpipe as when confined to the 2 sq. in.
standpipe. Yet the volume of water, or "charge"
is the same in either case.
Likewise a 1 pfd. capacitor charged to 1000
volts possesses twice as much potential
energy as does a 2 pfd. capacitor charged to
500 volts, though the charge (expressed in
coulombs: Q = CE) is the same in either case.

a

micro-microlarad = one - millionth of one millionth of a farad, or 10'E' farads.

Stored energy in joules

of the electrons would act on o free electron
located between the two capacitor plates.

micro farad = 1 /1,000,000 of
.000001 farad, or 10-6 farads.

micro- microfarad = 1 /1,000,000 of a micro farad, or .000001 microfarad, or 10'6 micro farads.

If the capacitance is to be expressed in
microfarads in the equation given for energy
storage, the factor C would then have to be
divided by 1,000,000, thus:

CHARGING CURRENT

1

31

or

Although any substance which has
the characteristics of a good insulator may be used as a dielectric material, commercially manufactured capacitors make use of dielectric materials
which have been selected because their characteristics are particularly suited to the job at
hand. Air is a very good dielectric material,
but an air - spaced capacitor does not have a
high capacitance since the dielectric constant
of air is only slightly greater than one. A
group of other commonly used dielectric mate ials is listed in figure 15.
Certain materials, such as bakelite, lucite,
and other plastics dissipate considerable
energy when used as capacitor dielectrics.
Dielectric
Materials

1

DIELECTRIC
CONSTANT

MATERIAL

1O

ANILINE- FORMALDEHYDE
RESIN
BARIUM TITANATE

MC.

POWER

FACTOR
1O

MC.

3 4

0.004

1200

1.0

.67

CASTOR OIL

3.7

CELLULOSE ACETATE
GLASS.WINDOW

6

GLASS, PYREX
FLUOROTHENE
XEL -F
METHYL - METHACRYLATE
LUCITE
MICA
MYCALEX, MYKROY
PHENOL -FORMALDEHYDE,
LOW-LOSS YELLOW
PHENOL -FORMALDEHYDE
BLACK BAKELITE
PORCELAIN
POLYETHYLENE
POLYSTYRENE
QUARTZ FUSED
RUBBER, HARD-EBONITE
STEATITE
SULFUR
TEFLON
TITANIUM DIOXIDE
TRANSFORMER OIL
UREA -FORMALDEHYDE
VINYL RESINS
WOOD. MAPLE

-

-6

0.04
POOR

SOFTENING
POINT
FAHRENHEIT

260

-

IRO

2000

4.5

0.02

U.S

0.6

-

2.6
5.4
7.0
5.0

0.007

160

5.5

0.03

7.0
25

0.005
0.0003
0.0002
0 0002
0.007
0.003
0.003
0.02
0.0006
0.003
0.05
0.02

2

2.55
4.2
2.6
6.1

3.6
2.1
100 -175

2.2
5.0
4.0

.

FIGURE 15

0.0003

0.002
0.015

POOR

650

270
330
_2600

220
175'

2600
150
2700'

236

-

2700

260
200

Direct Current Circuits

34

1

C

1

1

1

C,

1

1

1

1

C,

C,

C,

C,

capacitor is connected into a direct -current circuit, it will block
the d.c., or stop the flow of current. Beyond
the initial movement of electrons during the
period when the capacitor is being charged,
there will be no flow of current because the
circuit is effectively broken by the dielectric
of the capacitor.
Strictly speaking, a very small current may
actually flow because the dielectric of the
capacitor may not be a perfect insulator. This
minute current flow is the leakage current
previously referred to and is dependent upon
the internal d -c resistance of the capacitor.
This leakage current is usually quite noticeable in most types of electrolytic capacitors.
When an alternating current is applied to
a capacitor, the capacitor will charge and discharge a certain number of times per second
in accordance with the frequency of the alternating voltage. The electron flow in the charge
and discharge of a capacitor when an a-c
potential is applied constitutes an alternating
current, in effect. It is for this reason that a
capacitor will pass an alternating current yet
offer practically infinite opposition to a direct
current. These two properties are repeatedly
in evidence in a radio circuit.
Capacitors in
and

D -C

EQUAL
RESISTANCE

EQUAL
CAPACITANCE

When a

A -C

Circuits

good paper dielectric
filter capacitor has such a
high internal resistance (inin Series
dicating a good dielectric)
that the exact resistance will vary considerably from capacitor to capacitor even though
they are made by the same manufacturer and
are of the same rating. Thus, when 1000 volts
d.c. is connected across two 1-pfd. 500-volt
capacitors in series, the chances are that the
voltage will divide unevenly and one capacitor
will receive more than 500 volts and the other
less than 500 volts.
Voltage Rating
of Capacitors

RADIO

1

- +- - +- - +C,

THE

Any

connecting a half 1 -watt carbon resistor across each capacitor, the voltage will be equalized because the
resistors act as a voltage divider, and the
internal resistances of the capacitors are so
much higher (many megohms) that they have
but little effect in disturbing the voltage divider balance.
Carbon resistors of the inexpensive type
are not particularly accurate (not being designed for precision service); therefore it is
Voltage Equalizing
Resistors

By

megohm

Figure

18

SHOWING THE USE OF VOLTAGE EQUALIZING RESISTORS ACROSS CAPACITORS
CONNECTED IN SERIES

advisable to check several
ohmmeter to find two that
possible in resistance. The
is unimportant, just so it is
two resistors used.

on an accurate

are as close as
exact resistance
the same for the

capacitors are connected in series, alternating
voltage pays no heed to the
relatively high internal resistance of each
capacitor, but divides across the capacitors
in inverse proportion to the capacitance. Because, in addition to the d.c. across a capacitor in a filter or audio amplifier circuit there
is usually an a -c or a -f voltage component, it
is inadvisable to series -connect capacitors
of unequal capacitance even if dividers are
provided to keep the d.c. within the ratings of
the individual capacitors.
For instance, if a 500 -volt 1 -µfd. capacitor
is used in series with a 4-pfd. 500 -volt capacitor across a 250 -volt a -c supply, the 1 -µfd.
capacitor will have 200 volts a.c. across it
and the 4-pfd. capacitor only 50 volts. An
equalizing divider to do any good in this case
would have to be of very low resistance because of the comparatively low impedance of
the capacitors to a.c. Such a divider would
draw excessive current and be impracticable.
The safest rule to follow is to use only
capacitors of the same capacitance and voltage rating and to install matched high resistance proportioning resistors across the various
capacitors to equalize the d-c voltage drop
across each capacitor. This holds regardless
of how many capacitors are series -connected.
Capacitors in
Series on A.C.

When two

Electrolytic capacitors use a very
thin film of oxide as the dielectric, and are polarized; that is,
they have a positive and a negative terminal
which must be properly connected in a circuit;
otherwise, the oxide will break down and the
capacitor will overheat. The unit then will no
longer be of service. When electrolytic capacitors are connected in series, the positive terminal is always connected to the positive lead
of the power supply; the negative terminal of
Electrolytic
Capacitors

HANDBOOK

M

the capacitor connects to the positive terminal
of the next capacitor in the series combination.
The method of connection for electrolytic capacitors in series is shown in figure 18. Electrolytic capacitors have very low cost per
microfarad of capacity, but also have a large
power factor and high leakage; both dependent
upon applied voltage, temperature and the age
of the capacitor. The modern electrolytic capacitor uses a dry paste electrolyte embedded
in a gauze or paper dielectric. Aluminium foil
and the dielectric are wrapped in a circular
bundle and are mounted in a cardboard or metal
box. Etched electrodes may be employed to
increase the effective anode area, and the
total capacity of the unit.
The capacity of an electrolytic capacitor is
affected by the applied voltage, the usage of
the capacitor, and the temperature and humidity
of the environment. The capacity usually drops
with the aging of the unit. The leakage current
and power factor increase with age. At high
frequencies the power factor becomes so poor
that the electrolytic capacitor acts as a series
resistance rather than as a capacity.
Magnetism

2 -4

and Electromagnetism

The common bar or horseshoe magnet is
familiar to most people. The magnetic field
which surrounds it causes the magnet to attract other magnetic materials, such as iron
nails or tacks. Exactly the same kind of magnetic field is set up around any conductor
carrying a current, but the field exists only
while the current is flowing.
Magnetic Fields

Before a potential, or voltage, is applied to a con-

ductor there is no external field, because there
is no general movement of the electrons in
one direction. However, the electrons do progressively move along the conductor when an
e.m.f. is applied, the direction of motion depending upon the polarity of the e.m.f. Since
each electron has an electric field about it, the
flow of electrons causes these fields to build
up into a resultant external field which acts in
a plane at right angles to the direction in
which the current is flowing. This field is
known as the magnetic field.
The magnetic field around a current-carrying
conductor is illustrated in figure 19. The
direction of this magnetic field depends entirely upon the direction of electron drift or
current flow in the conductor. When the flow
is toward the observer, the field about the
conductor is clockwise; when the flow is away
from the observer, the field is counter- clockwise. This is easily remembered if the left
hand is clenched, with the thumb outstretched

agnetism

35

ELECTRON DRIFT
.."-SWITCH

Figure

19

LEFT -HAND RULE
Showing the direction of the magnetic lines of
force produced around a conductor carrying an
electric current.

and pointing in the direction of electron flow.
The fingers then indicate the direction of the
magnetic field around the conductor.
Each electron adds its field to the total external magnetic field, so that the greater the
number of electrons moving along the conductor, the stronger will be the resulting field.
One of the fundamental laws of magnetism
is that like poles repel one another and unlike
poles attract one another. This is true of current- carrying conductors as well as of permanent magnets. Thus, if two conductors are placed
side by side and the current in each is flowing
in the same direction, the magnetic fields will
also be in the same direction and will combine
to form a larger and stronger field. If the current flow in adjacent conductors is in opposite
directions, the magnetic fields oppose each
other and tend to cancel.
The magnetic field around a conductor may
be considerably increased in strength by winding the wire into a coil. The field around each
wire then combines with those of the adjacent
turns to form a total field through the coil
which is concentrated along the axis of the
coil and behaves externally in a way similar
to the field of a bar magnet.
If the left hand is held so that the thumb
is outstretched and parallel to the axis of a
coil, with the fingers curled to indicate the
direction of electron flow around the turns of
the coil, the thumb then points in the direction of the north pole of the magnetic field.
The Magnetic

In the magnetic circuit, the
units which correspond to current, voltage, and resistance
in the electrical circuit are flux, magneto motive force, and reluctance.

Circuit

Flux, Flux
Density

is made up of a drift
of electrons, so is a magnetic
field made up of lines of force, and
the total number of lines of force in a given
magnetic circuit is termed the flux. The flux
depends upon the material, cross section, and
length of the magnetic circuit, and it varies
directly as the current flowing in the circuit.
As a current

www.americanradiohistory.com

Direct Current Circuits

36

The unit of flux is the maxwell, and the symbol is the Greek letter cp (phi).
Flux density is the number of lines of force
per unit area. It is expressed in gauss if the
unit of area is the square centimeter (1 gauss
= 1 line of force per square centimeter), or
in lines per square inch. The symbol for flux
density is B if it is expressed in gausses, or
B if expressed in lines per square Inch.
The force which produces a
flux in a magnetic circuit
is called magnetomotive force.
It is abbreviated m.m.f. and is designated by
the letter F. The unit of magnetomotive force
is the gilbert, which is equivalent to 1.26 x NI,
where N is the number of turns and I is the
current flowing in the circuit in amperes.
The m.m.f. necessary to produce a given
flux density is stated in gilberts per centimeter (oersteds) (H), or in ampere -turns per
inch (H).
magnetomotive
Force

Magnetic reluctance corresponds
to electrical resistance, and is
the property of a material that opposes the
creation of a magnetic flux in the material.
It is expressed in rels, and the symbol is the
letter R. A material has a reluctance of 1 rel
when an m.m.f. of 1 ampere -turn (NI) generates
a flux of 1 line of force in it. Combinations
of reluctances are treated the same as resistances in finding the total effective reluctance. The specific reluctance of any substance is its reluctance per unit volume.
Except for iron and its alloys, most common
materials have a specific reluctance very
nearly the same as that of a vacuum, which,
for all practical purposes, may be considered
the same as the specific reluctance of air.
Reluctance

Ohm's Law for
The relations between flux,
Magnetic Circuits magnetomotive force, and

reluctance are exactly the
same as the relations between current, voltage, and resistance in the electrical circuit.
These can be stated as follows:
F

F
R

THE

duce in air. It may be expressed by the ratio
B/H or B/H. In other words,
B

ç

= flux, F =

B

or

R

H

H

where p is the premeability, B is the flux
density in gausses, B is the flux density in
lines per square inch, H is the m.m.f. in
gilberts per centimeter (oersteds), and H is
the m.m.f. in ampere -turns per inch. These
relations may also be stated as follows:
B

H=-

or

fi

B
H=-,
f

and B=Hit or

B=

Permeability is similar to electric
conductivity. There is, however,
one important difference: the permeability of
magnetic materials is not independent of the
magnetic current (flux) flowing through it,
although electrical conductivity is substantially independent of the electric current in a
wire. When the flux density of a magnetic
conductor has been increased to the saturation
point, a further increase in the magnetizing
force will not produce a corresponding increase in flux density.
Saturation

magnetic circuit
a magnetization
curve may be drawn for a given unit of material. Such a curve is termed a B -H curve, and
may be determined by experiment. When the
current in an iron core coil is first applied,
the relation between the winding current and
the core flux is shown at A -B in figure 20. If
the current is then reduced to zero, reversed,
brought back again to zero and reversed to the
Calculations

To

simplify

calculations,

F=chR

-

MAGNETIZING FORCE

m.m.f., and

R =

H

reluctance.

Permeability expresses the ease
with which a magnetic field may
be set up in a material as compared with the
effort required in the case of air. Iron, for example, has a permeability of around 2000
times that of air, which means that a given
amount of magnetizing effect produced in an
iron core by a current flowing through a coil
of wire will produce 2000 times the flux density
that the same magnetizing effect would pro-

Hµ

It can be seen from the foregoing that permeability is inversely proportional to the
specific reluctance of a material.

R

where

RADIO

Permeability

Figure

20

TYPICAL HYSTERESIS LOOP
(B -H CURVE = A -B)
Showing relationship between the current in the
winding of on iron core inductor and the core
Inducflux. A direct current flowing through
tance brings the magnetic state of the core to
some point on the hysteresis loop, such as C.

www.americanradiohistory.com

th

Inductance

HANDBOOK
original direction, the flux passes through a
typical hysteresis loop as shown.
The magnetism remaining
in a material after the
magnetizing force is removed is called residual magnetism. Retentivity is the property which causes a magnetic
material to have residual magnetism after
having been magnetized.
Residual Magnetism;

Retentivity

Hysteresis;
Coercive Force

Hysteresis is the character -

istic of a magnetic system
which causes a loss of power

due to the fact that a negative magnetizing
force must be applied to reduce the residual
magnetism to zero. This negative force is
termed coercive /orce. By "negative" magnetizing force is meant one which is of the
opposite polarity with respect to the original
magnetizing force. Hysteresis loss is apparent
in transformers and chokes by the heating of
the core.

the current. Thus, it can be seen that selfinduction tends to prevent any change in the
current in the circuit.
The storage of energy in a magnetic field
is expressed in joules and is equal to (LI3) /2.
(A joule is equal to 1 watt- second. L is defined immediately following.)

Inductance is usually denoted by
the letter L, and is expressed in
henrys. A coil has an inductance
of 1 henry when a voltage of 1
volt is induced by a current change of 1 ampere per second. The henry, while commonly
used in audio frequency circuits, is too large
for reference to inductance coils, such as
those used in radio frequency circuits; millihenry or microhenry is more commonly used,
in the following manner:
The Unit of
Inductance;
The Henry

1

1

If the switch shown in figure 19
is opened and closed, a pulsating
direct current will be produced. When it is
first closed, the current does not instantaneously rise to its maximum value, but builds
up to it. While it is building up, the magnetic
field is expanding around the conductor. Of
course, this happens in a small fraction of a
second. If the switch is then opened. the current stops and the magnetic field contracts
quickly. This expanding and contracting field
will induce a current in any other conductor
that is part of a continuous circuit which it
cuts. Such a field can be obtained in the way
just mentioned by means of a vibrator inter ruptor, or by applying a.c. to the circuit in
place of the battery. Varying the resistance of
the circuit will also produce the same effect.
This inducing of a current in a conductor due
to a varying current in another conductor not
in acutal contact is called electromagnetic induction.
Inductance

Self -inductance

If an alternating current flows
through a coil the varying
magnetic field around each turn cuts itself and
the adjacent turn and induces a voltage in the
coil of opposite polarity to the applied e.m.f.
The amount of induced voltage depends upon
the number of turns in the coil, the current
flowing in the coil, and the number of lines
of force threading the coil. The voltage so
induced is known as a counter-e.m. f. or back e.m.f., and the effect is termed self -induction.
When the applied voltage is building up, the
counter- e.m.f. opposes the rise; when the applied voltage is decreasing, the counter- e.m.f.
is of the same polarity and tends to maintain

37

henry = 1,000
henrys.

1

or

10'

milli -

millihenry = 1 /1,000 of a henry, .001 henry,
or

1

millihenrys,

10'

henry.

microhenry = 1 /1,000,000 of a henry,
.000001 henry, or 10-e henry.

or

microhenry =1/1,000 of a millihenry, .001
or 10-' millibenrys.

1,000 microbenrys =

millihenry.

1

coil is near another, a varying current in
one will produce a varying magnetic field
which cuts the turns of the other coil, inducing
a current in it. This induced current is also
varying, and will therefore induce another current in the first coil. This reaction between
two coupled circuits is called mutual induction,
and can be calculated and expressed in henrys.
The symbol for mutual inductance is M. Two
circuits thus joined are said to be inductively
coupled.
The magnitude of the mutual inductance depends upon the shape and size of the two circuits, their positions and distances apart, and
the premeability of the medium. The extent to
When one

Mutual Inductance

i
I

i.,

u

I

I

2

I

Figure 21
MUTUAL INDUCTANCE
The quantity M represents the mutual inductance
between the two coils L1 and L,.

www.americanradiohistory.com

Direct Current Circuits

38

i--

L

---¡

INDUCTANCE OF
SINGLE- LAYER
SOLENOID COILS
R2 N2
9R +10 L

L

WHERE

R

=

L

=

RADIUS OF COIL
LENGTH OF COIL

N

=

NUMBER OF TURNS

MICRONENRIES

TO CENTER OF

WIRE

Figure 22
FORMULA FOR
CALCULATING INDUCTANCE
Through the usa of the equation and the sketch
shown above
inductance of single -layer
solenoid coils can be calculated with an accuracy of about on. per cent for tho types of
coils normally used in the h -f and v -h -f range.

th

which two inductors are coupled is expressed
by a relation known as coefficient of coupling.
This is the ratio of the mutual inductance actually present to the maximum possible value.
The formula for mutual inductance is L
L, + L, + 2M when the coils are poled so that
their fields add. When they are poled so that
their fields buck, then L = L, + L, - 2M
(figure 21).

Inductors in parallel are corn bined exactly as are resistors in
parallel, provided that they are
far enough apart so that the mutual inductance
is entirely negligible.
Inductors in

Parallel

Inductors in series are additive,
just as are resistors in series,
again provided that no mutual
inductance exists. In this case, the total inductance L is:
Inductors in

Series

L =

L,

etc.

+ L2 +

Where mutual inductance does exist:
L

=L, +L,+

M is the mutual inductance.
This latter expression assumes that the
coils are connected in such a way that all flux
linkages are in the same direction, i.e., additive. If this is not the case and the mutual
linkages subtract from the self -linkages, the
following formula holds:

L
M

=L,

+L,- 2M,

as the frequency is increased. The principal
use for conventional magnetic cores is in the
audio -frequency range below approximately
15,000 cycles, whereas at very low frequencies
(50 to 60 cycles) their use is mandatory if
an appreciable value of inductance is desired.
An air core inductor of only 1 henry inductance would be quite large in size, yet
values as high as 500 henrys are commonly
available in small iron core chokes. The inductance of a coil with a magnetic core will
vary with the amount of current (both a-c and
d-c) which passes through the coil. For this
reason, iron core chokes that are used in power
supplies have a certain inductance rating at a
predetermined value of d-c.
The premeability of air does not change
with flux density; so the inductance of iron
core coils often is made less dependent upon
flux density by making part of the magnetic
path air, instead of utilizing a closed loop of
iron. This incorporation of an air gap is necessary in many applications of iron core coils,
particularly where the coil carries a considerable d -c component. Because the permeability
of air is so much lower than that of iron, the
air gap need comprise only a small fraction of
the magnetic circuit in order to provide a substantial proportion of the total reluctance.
Iron Cored Inductors
at Radio Frequencies

Iron -core inductors may
be used at radio frequencies if the iron is in a
very finely divided form, as in the case of the
powdered iron cores used in some types of r -f
coils and i -f transformers. These cores are
made of extremely small particles of iron. The
particles are treated with an insulating material so that each particle will be insulated from

the others, and the treated powder is molded
with a binder into cores. Eddy current losses
are greatly reduced, with the result that these
special iron cores are entirely practical in circuits which operate up to 100 Mc. in frequency.

2 -5

and R L

R C

voltage divider may be constructed as
figure 23. Kirchhoff's and Ohm's
Laws hold for such a divider. This circuit is
known as an RC circuit.
A

-

Circuits

Ordinary magnetic cores cannot be used for radio frequencies because the eddy current and hysteresis
losses in the core material becomes enormous

Transients

shown in

Time Constant
RC and RL

is the mutual inductance.

Core Material

RADIO

2M,

where

where

THE

When switch S in figure 23 is
placed in position 1, a volt meter across capacitor C will
indicate the manner in which

the capacitor will become charged through the
resistor R from battery B. If relatively large
values are used for R and C, and if a v -t voltmeter which draws negligible current is used

www.americanradiohistory.com

HANDBOOK

Time

Constant

39

to measure the voltage e, the rate of charge of
the capacitor may actually be plotted with the
aid of a stop watch.

It will be found that the voltage e will begin to rise
rapidly from zero the instant the switch is
closed. Then, as the capacitor begins to
charge, the rate of change of voltage across
the capacitor will be found to decrease, the
charging taking place more and more slowly
as the capacitor voltage e approaches the battery voltage E. Actually, it will be found that
in any given interval a constant percentage of
the remaining difference between e and E
will be delivered to the capacitor as an increase in voltage. A voltage which changes in
this manner is said to increase logarithmically,
or is said to follow an exponential curve.

Voltage Gradient

t001
r

:

60

L<

60
W

V40
<

ti

Wo
rt

20

OTIME

44 100
<`
<
FaA
60
óóZ

t. IN TERMS

OF

TIME CONSTANT

PC'

030 60
Hi
Iug
4.9

40.

ózó

0

óZW

<7

á 7ló1
ñ51

20¡C

- --

0

TIME

Time Constant

-

t, IN TERMS OF TIME
Figure

2

A

mathematical

analysis of

the charging of a capacitor in
this manner would show that the relationship
between the battery voltage E and the voltage
across the capacitor e could be expressed in
the following manner:

13

CONSTANT RC

23

TIME CONSTANT OF AN R -C CIRCUIT
Shown at (A) is the circuit upon which is based
the curves of (B) and (C). (8) shows the rate at
which capacitor C will charge from the instant
at which switch S is placed in position 1. (C)
shows the discharge curve of capacitor C from
the instant at which switch S is placed in
position 3.

e = E (1 _ f

-t /Rc)

where e,E,R, and C have the values discussed
above. f = 2.716 (the base of Naperian or
natural logarithms), and t represents the time
which has elapsed since the closing of the
switch. With t expressed in seconds, R and C
Figure

24

TYPICAL INDUCTANCES
The large inductance is a 1000 -watt transmitting coil. To the right and left of this coil are small r -f
chokes. S
I varieties of low power capability coils are shown below, along with various types of r -f
chokes intended for high- frequency operation.

www.americanradiohistory.com

Direct Current Circuits

40

R

(INCLUDING D.C. RESISTANCE
OF INDUCTOR

L)

i4cc

means that the voltage across the capacitor will have increased to 63.2 per cent of
the battery voltage in an interval equal to the
time constant or RC product of the circuit.
Then, during the next period equal to the time
constant of the RC combination, the voltage
across the capacitor will have risen to 63.2
per cent of the remaining difference in voltage,
or 86.5 per cent of the applied voltage E.
RL Circuit

TIME

t, IN TERMS OF

TIME CONSTANT

}

Figure 25

In the case of

a series combination
of a resistor and an inductor, as
shown in figure 25, the current through the
combination follows a very similar law to that
given above for the voltage appearing across
the capacitor in an RC series circuit. The
equation for the current through the combination is:

TIME CONSTANT OF AN R -L CIRCUIT
Nota that the time constant for the Increase In
current through an R-L. circuit Is identical to
the

rate of Increase in voltage across the
capacitor In on R -C circuit.

may be expressed in farads and ohms, or R
and C may be expressed in microfarads and
megohms. The product RC is called the time
constant of the circuit, and is expressed in
seconds. As an example, if R is one megohm
and C is one microfarad, the time constant
RC will be equal to the product of the two,
or one second.
When the elapsed time t is equal to the
time constant of the RC network under consideration, the exponent of E becomes -1.
Now
is equal to 1 /e, or 1/2.716, which
is 0.368. The quantity (1 - 0.368) then is equal
to 0.632. Expressed as percentage, the above

e'

i=-E

(1-E-tR/L)

where i represents the current at any instant
through the series circuit, E represents the
applied voltage, and R represents the total
resistance of the resistor and the d-c resistance of the inductor in series. Thus the time
constant of the RL circuit is L /R, with R expressed in ohms and L expressed in henrys.
Voltage Decoy

When the

switch in figure

23

is

moved to position 3 after the
capacitor has been charged, the capacitor voltage will drop in the manner shown in figure
23 -C. In this case the voltage across the capacitor will decrease to 36.8 per cent of the
initial voltage (will make 63.2 per cent of the
total drop) in a period of time equal to the
time constant of the RC circuit.

TYPICAL IRON -CORE INDUCTANCES
At the right is an upright mounting filter choke intended for use in low powered transmitters and audio equipment. At the center is o hermetically sealed inductance for use
under poor environmental conditions. To the left is an inexpensive receiving -type choke,
with a small iron -core r -f choke directly in front of it.

www.americanradiohistory.com

CHAPTER THREE

Alternating Current Circuits

present the usable frequency range for alternating electrical currents extends over the enormous frequency range from about 15 cycles per
second to perhaps 30,000,000,000 cycles per
second. It is obviously cumbersome to use a
frequency designation in c.p.s. for enormously
high frequencies, so three common units which
are multiples of one cycle per second have
been established.

The previous chapter has been devoted to
discussion of circuits and circuit elements
upon which is impressed a current consisting
of a flow of electrons in one direction. This
type of unidirectional current flow is called
direct current, abbreviated d. c. Equally as important in radio and communications work,
and power practice, is a type of current flow
whose direction of electron flow reverses
periodically. The reversal of flow may take
place at a low rate, in the case of power systems, or it may take place millions of times
per second in the case of communications
frequencies. This type of current flow is
called alternating current, abbreviated a. c.

At

Frequency Spectrum

a

z

4-

Y.1

¢

K
U

3 -1

TIME-41.

a

DIRECT CURRENT

Alternating Current

t CYCLE

Frequency of on
Alternating Current

An

-i

alternating current is

one whose amplitude of
current flow periodically
rises from zero to a maximum in one direction,
decreases to zero, changes its direction,
rises to maximum in the opposite direction,
and decreases to zero again. This complete
process, starting from zero, passing through
two maximums in opposite directions, and returning to zero again, is called a cycle. The
number of times per second that a current
passes through the complete cycle is called
the frequency of the current. One and one
quarter cycles of an alternating current wave
are illustrated diagrammatically in figure 1.

Iz
w

CYCLE

-01

TIME

a

CC

J

U

ALTERNATING CURRENT

Figure

1

ALTERNATING CURRENT
AND DIRECT CURRENT
Graphical comparison between unidrectionai
(direct) current and alternating current as plotted
against time.

41

www.americanradiohistory.com

-

42

Alternating Current Circuits

THE

RADIO

These units are:
(1) the kilocycle (abbr., kc.), 1000 c.p.s.
(2) the Megacycle (abbr., Mc.), 1,000,000
c.p.s. or 1000 kc.
(3) the kilo -Megacycle (abbr., kN1c.),
1,000,000,000 c.p.s. or 1000 Mc.

easily handled units such as these we
can classify the entire usable frequency range
into frequency bands.
The frequencies falling between about 15
and 20,000 c.p.s. are called audio frequencies,
abbreviated a.f., since these frequencies are
audible to the human ear when converted from
electrical to acoustical signals by a loudspeaker or headphone. Frequencies in the
vicinity of 60 c.p.s. also are called power frequencies, since they are commonly used to
distribute electrical power to the consumer.
The frequencies falling between 10,000
c.p.s. (10 kc.) and 30,000,000.000 c.p.s. (30
kMc.) are commonly called radio frequencies,
abbreviated r. J., since they are commonly used
in radio communication and allied arts. The
radio- frequency spectrum is often arbitrarily
classified into seven frequency bands, each
one of which is ten times as high in frequency
as the one just below it in the spectrum (except for the v -1 -f band at the bottom end of
the spectrum). The present spectrum, with
classifications, is given below.
With

Frequency
kc.
30 to 300 kc.
300 to 3000 kc.
3 to 30 Mc.
30 to 300 Mc.
300 to 3000 Mc.
3 to 30 kMc.
30 to 300 kMc.
10 to 30

Generation of
Alternating Current

Classification
Very -low frequencies
Low frequencies
Medium frequencies
High frequencies
Very -high frequencies
Ultra -high frequencies

Abbrev.

v.l.f
l.f.
m.f.
h. f.

v.h.f.
u.h.f.
Super-high frequencies s.h.f.
Extremely -high
frequencies
e.h.f.
Faraday discovered that if

a conductor which forms
part of a closed circuit is
moved through a magnetic field so as to cut
across the lines of force, a current will flow
in the conductor. He also discovered that, if a
conductor in a second closed circuit is brought
near the first conductor and the current in the
first one is varied, a current will flow in the
second conductor. This effect is known as
induction, and the currents so generated are
induced currents. In the latter case it is the

lines of force which are moving and cutting
the second conductor, due to the varying current strength in the first conductor.
A current is induced in a conductor if there
is a relative motion between the conductor
and a magnetic field, its direction of flow depending upon the direction of the relative

Figure 2
THE ALTERNATOR
Semi -schematic representation of the simplest
form of the alternator.

motion between the conductor and the field,
and its strength depends upon the intensity of
the field, the rate of cutting lines of force, and
the number of turns in the conductor.

machine that generates an alternating current is called an alternator or a -c generator. Such a machine in its
basic form is shown in figure 2. It consists of
two permanent magnets, M. the opposite poles
of which face each other and are machined so
that they have a common radius. Between
these two poles, north (N) and south (S),
a substantially constant magnetic field exists.
If a conductor in the form of C is suspended
so that it can be freely rotated between the
two poles, and if the opposite ends of conductor C are brought to collector rings, there
will be a flow of alternating current when conductor C is rotated. This current will flow out
through the collector rings R and brushes B
to the external circuit, X -Y.
The field intensity between the two pole
pieces is substantially constant over the entire
area of the pole face However, when the
conductor is moving parallel to the lines of
force at the top or bottom of the pole faces,
no lines are being cut. As the conductor moves
on across the pole face it cuts more and more
lines of force for each unit distance of travel,
until it is cutting the maximum number of
lines when opposite the center of the pole.
Therefore, zero current is induced in the conductor at the instant it is midway between
the two poles, and maximum current is induced when it is opposite the center of the
pole face. After the conductor has rotated
through 180° it can be seen that its position
with respect to the pole pieces will be exactly
opposite to that when it started. Hence, the
second 180° of rotation will produce an alternation of current in the opposite direction to
that of the first alternation.
The current does not increase directly as
the angle of rotation, but rather as the sine
of the angle; hence, such a current has the
mathematical form of a sine wave. Although
Alternators

A

www.americanradiohistory.com

HANDBOOK
LINES

Sine /lave

The

43

OF FORCE

t CYCLE
E

le2

90

60

A B C O E

CYCLE

30

HM-

--+

CYCLE- w

t20
ISO

te0
3

a
2

LINES OF FORCE
(UNIFORM DENSITY

240

2t0

3 CYCLE'

Graph showing sine -wave output current of the

alternator of figure

CYCLE' +-

WHERE

F =

TIMES

330

2143
t

Figure 3
OUTPUT OF THE ALTERNATOR

--

300

FREQUENCY IN CYCLES

2.

Figure

most electrical machinery does not produce a
strictly pure sine curve, the departures are
usually so slight that the assumption can be
regarded as fact for most practical purposes.
All that has been said in the foregoing paragraphs concerning alternating current also is
applicable to alternating voltage.
The rotating arrow to the left in figure 3
represents a conductor rotating in a constant
magnetic field of uniform density. The arrow
also can be taken as a vector representing the
strength of the magnetic field. This means that
the length of the arrow is determined by the
strength of the field (number of lines of force),
which is constant. Now if the arrow is rotating
at a constant rate (that is, with constant
angular velocity), then the voltage developed
across the conductor will be proportional to
the rate at which it is cutting lines of force,
which rate is proportional to the vertical
distance between the tip of the arrow and the

horizontal base line.
If EO is taken as unity or a voltage of 1,
then the voltage (vertical distance from tip of
arrow to the horizontal base line) at point C
for instance may be determined simply by
referring to a table of sines and looking up the
sine of the angle which the arrow makes with
the horizontal.
When the arrow has traveled from A to point
E, it has traveled 90 degrees or one quarter
cycle. The other three quadrants are not shown
because their complementary or mirror relationship to the first quadrant is obvious.
It is important to note that time units are
represented by degrees or quadrants. The fact
that AB, BC, CD, and DE are equal chords
(forming equal quadrants) simply means that
the arrow (conductor or vector) is traveling
at a constant speed, because these points on
the radius represent the passage of equal
units of time.
The whole picture can be represented in
another way, and its derivation from the foregoing is shown in figure 3. The time base is
represented by a straight line rather than by

4

THE SINE WAVE
illustrating

one cycle of o sine wave. One
complete cycle of alternation is broken up
into 360 degrees. Then one -half cycle is 180
degrees, one -quarter cycle is 90 degrees, and
so on down to the smallest division of the
wave. A cosine wave has a shape identical to
a sine wave but is shifted 90 degrees in phase
In other words the wove begfna at full am
plilude, the 90- degree point comes at zero amplitude, the 180 -degree point comes at full
amplitude in the opposite direction of current
How, etc.

-

angular rotation. Points A, B, C, etc., represent the same units of time as before. When
the voltage corresponding to each point is
projected to the corresponding time unit, the
familiar sine curve is the result.
The frequency of the generated voltage is
proportional to the speed of rotation of the
alternator, and to the number of magnetic poles
in the field. Alternators may be built to produce
radio frequencies up to 30 kilocycles, and
some such machines are still used for low
frequency communication purposes. By means
of multiple windings, three -phase output may
be obtained from large industrial alternators.
From figure 1 we see that the
value of an a -c wave varies
continuously. It is often of importance to know
the amplitude of the wave in terms of the
total amplitude at any instant or at any time
within the cycle. To be able to establish the
instant in question we must be able to divide
Radian Notation

the cycle into parts. We could divide the cycle
into eighths, hundredths, or any other ratio that
suited our fancy. However, it is much more
convenient mathematically to divide the cycle
either into electrical degrees (360° represent
one cycle) or into radians. A radian is an arc
of a circle equal to the radius of the circle;
hence there are 2n radians per cycle-or per
circle (since there are n diameters per circumference, there are 2rr radii).
Both radian notation and electrical degree

www.americanradiohistory.com

notation are used in discussions of alternating
current circuits. However, trigonometric tables
are much more readily available in terms of
degrees than radians, so the following simple
conversions are useful.
2n radians = 1 cycle = 3600
n radians ='/2 cycle = 180°

- radians ='4 cycle

RADIO

THE

Alternating Current Circuits

44

WHERE
o

e (THETA). PHASE

B

RADIANS

=

A

A

B'/r
D.
t

ANGLE

OR

RADIANS OR

T

.277F T

90'
IRO

RADIANS OR

2 A RADIANS OR

270
350

RADIAN a 57.324 DEGREES

n

=

2

- radians ='4 cycle

90°
Figure

n

=

60°

=

45°

3

-4 radians ='
n

/R

1

radian

cycle

Current

where e

The instantaneous volt age or current is proportional to the sine of the
angle through which the

rotating vector has travelled since reference
time t = 0. Hence, when the peak value of the
a -c wave amplitude (either voltage or current amplitude) is known, and the angle through
which the rotating vector has travelled is
established, the amplitude of the wave at
this instant can be determined through use
of the following expression:
e = Erna: sin

matical relationships involving phase angles
since such relationships are simplified when
radian notation is used

-cycle = 57.3°

When the conductor in the simple alternator of figure 2 has made one complete revolution it has generated one cycle and has rotated through 2n radians. The expression 2nf
then represents the number of radians in one
cycle multiplied by the number of cycles per
second (the frequency) of the alternating
voltage or current. The expression then represents the number of radians per second through
which the conductor has rotated. Hence 27rf
represents the angular velocity of the rotating
conductor, or of the rotating vector which
represents any alternating current or voltage,
expressed in radians per second.
In technical literature the expression 2nf
is often replaced by al, the lower -case Greek
letter omega. Velocity multiplied by time
gives the distance travelled. so 2nft (or an)
represents the angular distance through which
the rotating conductor or the rotating vector
has travelled since the reference time t = 0.
In the case of a sine wave the reference time
t = 0 represents that instant when the voltage
or the current, whichever is under discussion,
also is equal to zero.
Instantaneous Value

The radian is a unit of phase angle, equal to
57.324 degrees. It is commonly used in mathe-

1

=

2n

of Voltage or

5

ILLUSTRATING RADIAN NOTATION

left,

=

the instantaneous voltage

crest value of voltage,

E = maximum
f =

frequency in cycles per second, and

has elapsed
expressed as a fraction
of one second.

t = period of time which

since

t = 0

The instantaneous current can be found from
the same expression by substituting i for e

and Imax for Emaz.
It is often easier to visualize the process of

determining the instantaneous amplitude by
ignoring the frequency and considering only
one cycle of the a -c wave. In this case, for a
sine wave, the expression becomes:

e=Ema: sin 9
where O represents the angle through which
the vector has rotated since time (and amplitude) were zero. As examples:
when 0 = 30°
sin O = 0.5
so e = 0.5 Emu
when

O

=

sin O
so e

=
=

60°
0.866
0.866

Erna:

when

O = 90°
sin 0 = 1.0
so e = Emax

when

sin
so

www.americanradiohistory.com

O

=

=
e =
O

1 radian
0.8415
0.8415 Etna:

H

A-C

A N D B O O K

Effective Value

The instantaneous

of an

of an alternating current
or voltage varies continuously throughout the cycle.

Alternating Current

Relationships

45

value

value of an a -c wave must be chosen
to establish a relationship between the effectiveness of an a -c and a d -c voltage or cur rent/ The heating value of an alternating
current has been chosen to establish the reference between the effective values of a.c. and
d.c. Thus an alternating current will have an
effective value of 1 ampere when it produces
the same heat in a resistor as does 1 ampere
of direct current.
The effective value is derived by taking the
instantaneous values of current over a cycle of
alternating current, squaring these values.
taking an average of the squares, and then
taking the square root of the average. By this
procedure, the effective value becomes known
as the root mean square or r.m.s. value. This
is the value that is read on a -c voltmeters and
a -c ammeters. The r.m.s. value is 70.7 (for
sine waves only) per cent of the peak or maximum instantaneous value and is expressed as
So some

follows:
Eetf. or Er.m.s.

=

0.707 x

left. or Ir.m.s.

=

0.707 x !ma:.

Erna:

or

The following relations are extremely useful
in radio and power work:
Er

m. s. =

Ems

0.707 x

= 1.414 x

&max,

and

Er.m.s.

If

an alternating current
is passed through a rectifier, it emerges in the
form of a current of
varying amplitude which flows in one direction only. Such a current is known as rectified
a. c. or pulsating d. c. A typical wave form of a
pulsating direct current as would be obtained
from the output of a full -wave rectifier is
shown in figure 6.
Measuring instruments designed for d -c
operation will not read the peak for instantaneous maximum value of the pulsating d -c output from the rectifier; they will read only the
average value. This can be explained by assuming that it could be possible to cut off
some of the peaks of the waves, using the cutoff portions to fill in the spaces that are open,
thereby obtaining an average d -c value. A
milliammeter and voltmeter connected to the
adjoining circuit, or across the output of the
rectifier, will read this average value. It is related to peak value by the following expres-

Rectified Alternating
Current or Pulsating Direct Current

sion:
Eavg =

0.636 x Fina:

Figure 6
FULL -WAVE RECTIFIED
SINE WAVE
Waveform obtained at the output of a fullwave
rectifier being fed with a sine wave and having
100 per
cent rectification efficiency. Each
pulse has the same shape os one -half cycle of
a

sine wave. This type of current is known as
pulsating direct current.

It is thus seen that the average value is 63.6
per cent of the peak value.
To summarize the three
most significant values
of an a-c sine wave: the
Effective, and
Average Values
peak value is equal to
1.41 times the r.m.s. or
effective, and the r.m.s. value is equal to
0.707 times the peak value; the average value
of a full -wave rectified a-c wave is 0.636
times the peak value, and the average value
of a rectified wave is equal to 0.9 times the
r.m.s. value.
Relationship Between
Peak, R.M.S. or

= 0.707 x Peak
Average = 0.636 x Peak

R.M.S.

x R.M.S.
Average = 0.9
= 1.11 x Average
R.M.S.
Peak
Peak

= 1.414 x R.M.S.
= 1.57 x Average

law
applies
equally to direct or alternating current, provided the circuits under consideration are
purely resistive, that is, circuits which have
neither inductance (coils) nor capacitance
(capacitors). Problems which involve tube
filaments, drop resistors, electric lamps,
heaters or similar resistive devices can be
solved from Ohm's law, regardless of whether
the current is direct or alternating. When a
capacitor or coil is made a part of the circuit,
a property common to either, called reactance,
must be taken into consideration. Ohm's law
still applies to a -c circuits containing reactance, but additional considerations are involved; these will be discussed in a later
Applying Ohm's Law
to Alternating Current

paragraph.

www.americanradiohistory.com

Ohm's

Alternating Current Circuits

46

THE

RADIO

E

TIME

TIME

CURRENT LAGGING VOLTAGE BY 90°

CURRENT LEADING VOLTAGE BY 90°

(CIRCUIT CONTAINING PURE INDUCTANCE ONLY)

(CIRCUIT CONTAINING PURE CAPACITANCE ONLY)

Figure 7
LAGGING PHASE ANGLE

Figure 8
LEADING PHASE ANGLE

Showing the manner in which the current lags
the voltage in an a-c circuit containing pure
inductance only. The lag is equal to one -quarter
cycle or 90 degrees.

Inductive

As

Reactance

when

was stated in

Chapter Two,
changing current flows
through an inductor a back- or
counter -electromotive force is developed,
opposing any change in the initial current.
This property of an inductor causes it to offer
opposition or impedance to a change in current. The measure of impedance offered by an
inductor to an alternating current of a given
frequency is known as its inductive reactance.
This is expressed as XL.
a

XL

= 2rrf

L,

n= 3.1416 (2n= 6.283),
f = frequency in cycles,
L = inductance in henrys.
It is very often neces-

cary to compute inductive reactance at radio
frequencies. The same formula may be used,
but to make it less cumbersome the inductance
is expressed in millihenrys and the frequency
in kilocycles. For higher frequencies and
smaller values of inductance, frequency is
expressed in megacycles and inductance in
microhenrys. The basic equation need not be
changed, since the multiplying factors for
inductance and frequency appear in numerator
and denominator, and hence are cancelled out.
However, it is not possible in the same equation to express L in millihenrys and f in cycles

without conversion factors.
Capacitive
Reactance

Capacitors have a similar property although
in this case the opposition is to any change in
the voltage across the capacitor. This property
is called capacitive reactance and is expressed as follows:

Xc

It has been explained that inductive reactance is the measure of
the ability of an inductor to offer

impedance to the flow of an alternating current.

1

2nfC
where Xc = capacitive reactance in ohms,
n = 3.1416
f = frequency in cycles,
C =

where XL = inductive reactance expressed in
ohms.

Inductive Reactance
at Rodio Frequencies

Showing the manner in which the current leads
the voltage in on o -c circuit containing pure
capacitance only. The lead is equal to one quarter cycle or 90 degrees.

capacitance in farads.

Capacitive Reactance at
Radio Frequencies

Here again, as in the case
of inductive reactance, the
units of capacitance and

frequency can be converted

into smaller units for practical problems
en-

countered in radio work. The equation may
be

written:

1,000,000

Xc

2nfC

where f = frequency in megacycles,
C w capacitance in micro- microfarads.
In the audio range it is often convenient to
express frequency (f) in cycles and capacitance (C) in micro /arads, in which event the
same formula applies.
Phase

When

an

alternating current flows

through a purely resistive circuit, it
will be found that the current will go through
maximum and minimum in perfect step with
the voltage. In this case the current is said to
be in step or in phase with the voltage. For
this reason, Ohm's law will apply equally well
for a. c. or d. c. where pure resistances are concerned, provided that the same values of the

www.americanradiohistory.com

Reactance

HANDBOOK
Y-AniS

wave (either peak or r.m.s.) for both voltage
and current are used in the calculations.
However, in calculations involving alternating currents the voltage and current are not
necessarily in phase. The current through the
circuit may lag behind the voltage, in which
case the current is said to have lagging phase.

Lagging phase is caused by inductive reactance. If the current reaches its maximum value
ahead of the voltage (figure 8) the current is
said to have a leading phase. A leading phase
angle is caused by capacitive reactance.
In an electrical circuit containing reactance
only, the current will either lead or lag the
voltage by 90 °. If the circuit contains inductive reactance only, the current will lag the
voltage by 90 °. If only capacitive reactance is
in the circuit, the current will lead the voltage
by 90 °.

Inductive and capacitive reactance have exactly opposite
effects on the phase relation
between current and voltage in a circuit.
Hence when they are used in combination
their effects tend to neutralize. The combined
effect of a capacitive and an inductive reactance is often called the net reactance of a
circuit. The net reactance (X) is found by subtracting the capacitive reactance from the inductive reactance, X = XL Xc.
The result of such a combination of pure
reactances may be either positive, in which
case the positive reactance is greater so that
the net reactance is inductive, or it may be
negative in which case the capacitive reactance is greater so that the net reactance is
capacitive. The net reactance may also be
zero in which case the circuit is said to be
resonant. The condition of resonance will be
discussed in a later section. Note that inductive reactance is always taken as being positive while capacitive reactance is always
taken as being negative.

Reactances

in Combination

Pure reactances introduce a phase angle of
90° between voltage and
and Resistance
current; pure resistance
introduces no phase shift between voltage and
current. Hence we cannot add a reactance and
a resistance directly. When a reactance and a
resistance are used in combination the resulting phase angle of current flow with respect to the impressed voltage lies somewhere
between plus or minus 90° and 0° depending
upon the relative magnitudes of the reactance
and the resistance.
The term impedance is a general term which
can be applied to any electrical entity which
impedes the flow of current. Hence the term
may be used to designate a resistance, a pure
Impedance; Circuits
Containing Reactance

47

Figure

9

Operation on the vector (+A) by the quantity ( -1)
vector to rotate through 180 degrees.

reactance, or a complex combination of both
reactance and resistance. The designation for
impedance is Z. An impedance must be defined in such a manner that both its magnitude
and its phase angle are established. The
designation may be accomplished in either of
two ways-one of which is convertible into
the other by simple mathematical operations.

"J"

The first method of designating an impedance is
actually to specify both the resistive and the
reactive component in the form R + jX. In this
form R represents the resistive component in
ohms and X represents the reactive component.
The "j" merely means that the X component
is reactive and thus cannot be added directly
to the R component. Plus jX means that the
reactance is positive or inductive, while if
minus jX were given it would mean that the
reactive component was negative or capacitive.
In figure 9 we have a vector ( +A) lying along
the positive X-axis of the usual X -Y coordinate system. If this vector is multiplied by the
quantity ( -1), it becomes ( A) and its position
now lies along the X -axis in the negative
direction. The operator ( -1) has caused the
vector to rotate through an angle of 180 deThe

Operator

grees. Since ( -1) is equal to (V-1 x V77-1), the
same result may be obtained by operating on
the vector with the operator (VIT x V-1).
However if the vector is operated on but once
by the operator (V-1), it is caused to rotate
only 90 degrees (figure 10). Thus the operator
( V- 11)rotates a vector by 90 degrees. For convenience, this operator is called the j operator.
rotates the
In like fashion, the operator (
vector of figure 9 through an angle of 270
degrees, so that the resulting vector ( jA)
falls on the ( Y) axis of the coordinate system.

www.americanradiohistory.com

j)

48

Alternating Current Circuits

RADIO

THE

Y-AXIS

(AI

( +A) X
) ROTATES
VECTOR THROUGH 90

+jA

k

4

b

A

X

Z. 4+J3

AXIS

X

i
Figure

(j)

Polar Notation

The second method of representing an impedance is to
specify its absolute magnitude and the phase
angle of current with respect to voltage, in
the form Z L O. Figure 11 shows graphically
the relationship between the two common ways
of representing an impedance.
The construction of figure 11 is called an
impedance diagram. Through the use of such
a diagram we can add graphically a resistance
and a reactance to obtain a value for the resulting impedance in the scalar form. With
zero at the origin, resistances are plotted to
the right, positive values of reactance (inductive) in the upward direction, and negative
values of reactance (capacitive) in the downward direction.
Note that the resistance and reactance are
drawn as the two sides of a right triangle,
with the hypotenuse representing the resulting
impedance. Hence it is possible to determine mathematically the value of a resultant
impedance through the familiar right -triangle
relationship-the square of the hypotenuse is
equal to the sum of the squares of the other
two sides:

=R' +X2

or IZI = ß/R2 + X
Note also that the angle O included between
R and Z can be determined from any of the

following trigonometric relationships:
X

sin o

cos

tan

Z
R

O =

Z

X

O

=

-

IZI'

-R
OHMS

RESISTANCE

R.

5

IZI= 5

t0/-' 0.73
ae.e5

10

Operation on the vector ( +A) by the quantity
causes vector to rotate through 90 degrees.

Z2

o

L 3e.e5

R

One common problem is that of determining
the scalar magnitude of the impedance, IZI,

Figure 11
THE IMPEDANCE TRIANGLE
Showing the graphical construction of a triangle
for obtaining the net (scalar) impedance resulting from the connection of o resistance and
a reactance in series. Shown also alongside is
the
alternative mathematical procedure for
obtaining the values associated with the triangle.

and the phase angle 0, when resistance and
reactance are known; hence, of converting
from the Z = R + jX to the IZI LO form. In this
case we use two of the expressions just given:
IZI = V/R2+X2

tan

-, (or
X

O

=

O =

R

The

tan'

X

)

R

inverse problem,

that of converting
jX form is done
with the following relationships, both of which
are obtainable by simple division from the
trigonometric expressions just given for determining the angle 0:
from the IZI LO to the R +

R

jX

=IZI

cos0

=IZIj

sin

O

By simple addition these two expressions may
be combined to give the relationship between
the two most common methods of indicating
an impedance:
R +

jX =IZI (cos

B + j

sin 0)

In the case of impedance, resistance, or reactance, the unit of measurement is the ohm;
hence, the ohm may be thought of as a unit of
opposition to current flow, without reference
to the relative phase angle between the applied voltage and the current which flows.
Further, since both capacitive and inductive
reactance are functions of frequency, impedance will vary with frequency. Figure 12
shows the manner in which IZI will vary with
frequency in an RL series circuit and in an
RC series circuit.
Series RLC Circuits

www.americanradiohistory.com

In a series circuit containing R, L, and C, the im-

HANDBOOK

Impedance

49

pedance is determined as discussed before except that the reactive component in the expressions becomes: (The net reactance-the
difference between XL and Xc.) Hence (XL
Xc) may be substituted for X in the equations.

Thus:
IZI = VR' +(XL
O

= tan

(XL
'(XL

Xc)'
Xc

)

R

A series RLC circuit thus may present an
impedance which is capacitively reactive if
the net reactance is capacitive, inductively
reactive if the net reactance is inductive, or
resistive if the capacitive and inductive reactances are equal.

Addition of
Complex Quantities

The addition of complex
quantities (for example,
impedances in series) is
quite simple if the quantities are in the rectangular form. If they are in the polar form
they only can be added graphically, unless
they are converted to the rectangular form by
the relationships previously given. As an example of the addition of complex quantities
in the rectangular form, the equation for the
addition of impedances is:
( R,

+jX,) +(R, +jX,)= (R,

+ Rs)

+j(X,+X,)

For example if we wish to add the impedances (10 + j50) and (20 j30) we obtain:
(10 + j50) + (20
j30)
= (10 + 20) + j(50 + ( -30)
= 30 + j(50 -30)

It is often necessary in

solving certain types of
circuits to multiply or divide two complex quantities. It is a much simplier mathematical
operation to multiply or divide complex quantities if they are expressed in the polar form.
Hence if they are given in the rectangular
form they should be converted to the polar
form before multiplication or division is begun.
Then the multiplication is accomplished by
multiplying the IZ1 terms together and adding
algebraically the L e terms, as:

(IZ,1Le,)(I4,I Le,)

=IZ,1 Iz,I

L43°)

(1321

L -23 °)

approaches zero.

Division is accomplished by dividing the
denominator into the numerator, and subtracting the angle of the denominator from
that of the numerator, as:
IZ,I Le,

IZ,IL0,-

= 120.321
(L43° + L -23 °)
= 640 L 20°

IZ,I
1z21(Le,

tee,)

For example, suppose that an impedance of
1501 L 67° is to be divided by an impedance
of 1101 L45°. Then:
1501

1101

L67°
L45°

1501

= 151
1101

(L 22 °)

Ohm's Law for
Complex Quantities

The simple form of Ohm's
Law used for d -c circuits
may be stated in a more
general form for application to a -c circuits
involving either complex quantities or simple
resistive elements. The form is:
E

z

(L0,+ L0,)

For example, suppose that the two impedances
1201 L43
and 1321 L -23° are to be multiplied. Then:
( 1201

Figure 12
IMPEDANCE AGAINST FREQUENCY
FOR R L AND R -C CIRCUITS
The impedance of an R -C circuit approaches
infinity os the frequency approaches zero (d.c.),
while the impedance of o series R -L circuit
approaches infinity as the frequency approaches
infinity. The impedance of an R -C circuit approaches the impedance of the series resistor
os the frequency approaches infinity, while the
impedance of o series R -L circuit approaches
the impedance of the resistor as the frequency

)

= 30 + j20

Multiplication and
Division of
Complex Quantities

o

in which, in the general case, I, E, and Z are
complex (vector) quantities. In the simple
case where the impedance is a pure resistance
with an a -c voltage applied, the equation
simplifies to the familiar I = E /R. In any case
the applied voltage may be expressed either
as peak, r.m.s., or average; the resulting

www.americanradiohistory.com

Alternating Current Circuits

50

THE

Since the applied voltage will be the reference
for the currents and voltages within the circuit, we may define it as having a zero phase
angle: E = 100 LO °. Then:

200 n.

oo

_
Figure 13
R -L -C CIRCUIT

=

SERIES

current always will be in the same type of
units as used to define the voltage.
In the more general case vector algebra
must be used to solve the equation. And,
since either division or multiplication is involved, the complex quantities should be expressed in the polar form. As an example,
take the case of the series circuit shown in
figure 13 with 100 volts applied. The impedance of the series circuit can best be obtained
first in the rectangular form, as:

;(l00-.300) = 200
200 + j(100

V200'+(-200)'

= N/40,000 +
=

100 LO °

282 L -45°

40,000

This same current must flow through all three
elements of the circuit, since they are in
series and the current through one must already have passed through the other two.
Hence the voltage drop across the resistor
(whose phase angle of course is 0 °) is:
E

=

The voltage drop across the inductive reactance is:
E = I XL

(0.354 L45 °) (100 L 90 °)
= 35.4 L 135° volts
Similarly, the voltage drop across the capacitive reactance is:
E =

-

E = (0.354 1_45°) (300 /--90 °)
=

-200

X

tan' -=tan'

200

R

=

=

tan' -1

-45 °.
= 282

L -45°

Note that in a series circuit the resulting impedance takes the sign of the largest reactance in the series combination.
Where a slide-rule is being used to make
the computations, the impedance may be found
without any addition or subtraction operations
by finding the angle O first, and then using
the trigonometric equation below for obtaining the impedance. Thus:
O

-tan' -1
=tan'-XR =tan' -200
200
=

-45°
R

Then IZI

=

cos

cos -45°
O

200
IZI

L0°)

70.8 L 45° volts

E = I Xc

80,000

Therefore Z

=1R

E = (0.354 L 45 °) (200

= 282 f2

O=

-0 .354 L0 °- ( -45 °)

0.354 L45° amperes.

-j200

Now, to obtain the current we must convert
this impedance to the polar form.
IZI =

RADIO

0.707

=

0.707

Note that the voltage drop across the capacitive reactance is greater than the supply
voltage. This condition often occurs in a
series RLC circuit, and is explained by the
fact that the drop across the capacitive reactance is cancelled to a lesser or greater extent by the drop across the inductive reactance.
It is often desirable in a problem such as
the above to check the validity of the answer
by adding vectorially the voltage drops across
the components of the series circuit to make
sure that they add up to the supply voltage
or to use the terminology of Kirchhoff's Second
Law, to make sure that the voltage drops
across all elements of the circuit, including
the source taken as negative, is equal to zero.
In the general case of the addition of a
number of voltage vectors in series it is best
to resolve the voltages into their in -phase
and out-of -phase components with respect to
the supply voltage. Then these components
may be added directly. Hence:

-

ER =
=

- 28212

106.2 L -45°

=
=

70.8L45°
70.8 ( cos 45° + j sin 45 °)
70.8 (0.707 + j0.707)
50 + j50

www.americanradiohistory.com

HANDBOOK

Vector Algebra

51

ao

i

DROP ACROSS RESISTOR

,o

°'-

±leo

45

e

LI NE VOLTAGE =100

PARALLEL CIRCUIT

XL + XL .70.111/-45.

f0
Figure

=

=
=

Ec=

13.

35.4L135°
35.4 ( cos 135° + j sin 135 °)
35.4 ( -0.707 + j0.707)

-25

+ j25

106.2L45°

= 106.2 ( cos -45 °+ j sin -45 °)
= 106.2 (6.707 -j0.707)
= 75

-j75

=(50+ j50)
+(75 -j75)

ER + EL +EC

Figure

+

(-25

ments which go to make up the series circuit
is the same. But the voltage drops across
each of the components are, in general, different from one another. Conversely, in a
parallel RLC or RX circuit the voltage is,
obviously, the same across each of the elements. But the currents through each of the
elements are usually different.
There are many ways of solving a problem
involving paralleled resistance and reactance;
several of these ways will be described. In
general, it may be said that the impedance of
a number of elements in parallel is solved
using the same relations as are used for
solving resistors in parallel, except that complex quantities are employed. The basic re-

lation is:
+

j25)

1

Zrot
the

supply voltage.
It is frequently desirable
to check computations in-

volving complex quantities

by constructing vectors
representing the quantities on the complex
plane. Figure 14 shows such a construction
for the quantities of the problem just completed. Note that the answer to the problem
may be checked by constructing a parallelogram with the voltage drop across the resistor as one side and the net voltage drop
across the capacitor plus the inductor (these
may be added algrebraically as they are 180°
out of phase) as the adjacent side. The vector
sum of these two voltages, which is represented by the diagonal of the parallelogram,
is equal to the supply voltage of 100 volts at
zero phase angle.
Resistance and Reactonce in Parallel

-+
-+ -+
Z,

1

-25+

= (50
75) + j(50 + 25-75)
= 100 +j0
= 100 LO °, which is equal to

Checking by
Construction on the
Complex Plane

15

THE EQUIVALENT SERIES CIRCUIT
Showing a parallel R -C circuit and the equivalent series R -C circuit which represents the
same net impedance os the parallel circuit.

14

Graphical construction of the voltage drops
associated with the serles R -L -C circuit of

figure

EQUIVALENT SERIES CIRCUIT

-43

1

NET DROP ACROSS

,.n

-

.

DROP ACROSS XC =1Oe.2

EL =

T

:_e.i

VOLTAGE DROP ACROSS
X1.= 35.4
laa

series circuit, such
as just discussed, the current through all the ele-

1

1

Z2

Z,

or when only two impedances are involved:
Z, Z2
Z`o` Z t + Z :
As an example, using the two- impedance
relation, take the simple case, illustrated in
figure 15, of a resistance of 6 ohms in parallel with a capacitive reactance of 4 ohms. To
simplify the first step in the computation it is
best to put the impedances in the polar form
for the numerator, since multiplication is involved, and in the rectangular form for the

addition in the denominator.
Zrot

-

(6 L0°) (4

-90°)

6-j4
24
6

L-90°

-j4

Then the denominator is changed to the polar
form for the division operation:

In a

O

=

tañ'

-4

www.americanradiohistory.com

6

=

tan-'

- 0.667 = - 33.7°

THE

Alternating Current Circuits

52

6

IZI =

cos

7.21 ohms

0.832

33.7°

E.

j4 = 7.21 L-33.7°

6

°

Ztat

=

7.21

= 3.33

L
(

Rz
EzEi R1+Rz

L -90°
33.7 °=

cos

= 3.33 [ 0.5548 + j
= 1.85

j

3.33 L -56.3°

56.3° +

j

sin

(- 0.832)1

Through the series of operations in the previous
paragraph we have converted a circuit composed of two impedances in
parallel into an equivalent series circuit composed of impedances in series. An equivalent
series circuit is one which, as far as the terminals are concerned, acts identically to the
original parallel circuit; the current through
the circuit and the power dissipation of the
resistive elements are the same for a given
voltage at the specified frequency.
We can check the equivalent series circuit
of figure 15 with respect to the original circuit by assuming that one volt a.c. (at the
frequency where the capacitive reactance in
the parallel circuit is 4 ohms) is applied to
the terminals of both.
In the parallel circuit the current through
the resistor will be 2/6 ampere (0.166a.) while
the current through the capacitor will be j
ampere (+ j 0.25 a.). The total current will be
the sum of these two currents, or 0.166 +
j 0.25 a. Adding these vectorially we obtain:

W

1

=0.3a.
will

be:

=I2R =0.32x1.85
= 0.9 x 1.85
= 0.166 watts

that the equivalent series circuit
checks exactly with the original parallel circuit.
So we see

Parallel RLC
Circuits

In solving a more complicated

circuit

Ez-E

G+Cz

E2-E1

Lz

LI+Lz

Cz

O

elect to use either of two methods of
solution. These methods are called the admittance method and the assumed - voltage method.
However, the two methods are equivalent
since both use the sum-of-reciprocals equation:
may

1

Ztot

- +- +1

1

1

Z1

Z2

Zs

In the admittance method we use the relation
Y = 1 /Z, where Y = G + jB; Y is called the
admittance, defined above, G is the conductance or R /Z' and B is the susceptance or
X/Z2. Then Ytot = 1 /Ztot = Y1 + Y2 + Y,
In the assumed- voltage method we multiply
both sides of the equation above by E, the
assumed voltage, and add the currents, as:
E

Ztot

-+ -+- ...
E

E

E

Zt

Z,

Z3

= 1zt

+Iz2 +1zt

..

Then the impedance of the parallel combination may be determined from the relation:

Ztot = Eí IZ tot
Voltage dividers for use with
alternating current are quite similar to d-c voltage dividers. However, since capacitors and inductors oppose
the flow of a-c current as well as resistors,
voltage dividers for alternating voltages may
take any of the configurations shown in figDividers

The dissipation in the resistor will be 12/6 =
0.166 watts.
In the case of the equivalent series circuit
the current will be:

And the dissipation in the resistor

xo +502

AC Voltage

III = x/0.1662 + 0.252 = x/0.09 = 0.3 a.

3.33

xCz

Ez=E1

Figure 16
SIMPLE A -C VOLTAGE DIVIDERS

Equivalent Series
Circuit

E

+o

OA

56.3°)

2.77

Ill

,o
Ez

1

Then:
24

I

6

RADIO

made up of more than

two impedances in parallel we

ure 16.
Since the impedances within each divider
are of the same type, the output voltage is in
phase with the input voltage. By using com-

binations of different types of impedances, the
phase angle of the output may be shifted in
relation to the input phase angle at the same
time the amplitude is reduced. Several dividers of this type are shown in figure 17.
Note that the ratio of output voltage to input
voltage is equal to the ratio of the output
impedance to the total divider impedance.
This relationship is true only if negligible
current is drawn by a load on the output terminals.

www.americanradiohistory.com

HANDBOOK

Xc

E2Ei

Circuits

Resonant

E2

R2+XC2

E

53

XL

R2 +XL2

Figure

©

18

SERIES RESONANT CIRCUIT

If the values of inductance and capacitance
both are fixed, there will be only one resonant
E2E,

XL

Ez

XL-Xc
DO

E,

Ei

Ea

Es - Ei

R2+2
Xc

-

X
R2.2
XL-XCXC

R2+I (L-XC12

COMPLEX

3 -2

Figure 17
VOLTAGE DIVIDERS

A -C

Resonant Circuits

frequency.
If both the inductance and capacitance are
made variable, the circuit may then be changed
or tuned, so that a number of combinations
of inductance and capacitance can resonate at
the same frequency. This can be more easily
understood when one considers that inductive
reactance and capacitive reactance travel in
opposite directions as the frequency is changed.
For example, if the frequency were to remain
constant and the values of inductance and
capacitance were then changed, the following
combinations would have equal reactance:

Frequency is constant at 60 cycles.
L is expressed in henrys.

series circuit such as shown in figure 18
is said to be in resonance when the applied
frequency is such that the capacitive reactance is exactly balanced by the inductive reactance. At this frequency the two reactances
will cancel in their effects, and the impedance
of the circuit will be at a minimum so that
maximum current will flow. In fact, as shown
in figure 19 the net impedance of a series
circuit at resonance is equal to the resistance
which remains in the circuit after the reactances have been cancelled.
A

resistance is always
present in a circuit because it is possessed in some degree by both
the inductor and the capacitor. If the frequency of the alternator E is varied from
nearly zero to some high frequency, there will
be one particular frequency at which the inductive reactance and capacitive reactance
will be equal. This is known as the resonant
frequency, and in a series circuit it is the
frequency at which the circuit current will be
a maximum. Such series resonant circuits are
chiefly used when it is desirable to allow a
certain frequency to pass through the circuit
(low impedance to this frequency), while at
the same time the circuit is made to offer
considerable opposition to currents of other
frequencies.
R

C

is expressed in microfarads (.000001

farad.)
XL

L
.265
2.65
26.5
265.00
2,650.00

1,000
10,000
100,000
1,000,000

Frequency
of Resonance

100

1,000
10.000
100,000
1,000,000

From the formula for resonance, 2rrfL = 1 /2nfC. the resonant frequency is determined:

f=

t Frequency Some

Xc

C

26.5
2.65
.265
.0265
.00265

100

1

2rr

N/

LC

where f = frequency in cycles,

L = inductance in henrys,
C = capacitance in farads.

It is more convenient to express L and C
in smaller units, especially in making radio frequency calculations; f can also be expressed in megacycles or kilocycles. A very
useful group of such formulas is:

f2=

25,330

LC

orL=

25,330

f2C

orC=

25,330

f2L

where f = frequency in megacycles,
L = inductance in microhenrys,
C = capacitance in micromicrofarads.

www.americanradiohistory.com

54

Alternating Current Circuits

THE

Figure 19
IMPEDANCE OF A
SERIES -RESONANT CIRCUIT
Showing the variation in reactance of the separate elements and In the net impedance of o
series resonant circuit (such as figure 18) with
changing frequency. The vertical line is drawn
at the point of resonance (XL
Xc = 0) in the

-

series circuit.

Impedance of Series
Resonant Circuits
18)

is:

The impedance across
the terminals of a series
resonant circuit (figure

/r3

Z =
Xc)2,
+ (XL
where Z = impedance in ohms,
r = resistance in ohms,
Xc = capacitive reactance in ohms,
XL = inductive reactance in ohms.
From this equation, it can be seen that the
impedance is equal to the vector sum of the
circuit resistance and the difference between
the two reactances. Since at the resonant frequency XL equals Xc. the difference between
them (figure 19) is zero, so that at resonance
the impedance is simply equal to the resistance of the circuit; therefore, because the
resistance of most normal radio- frequency
circuits is of a very low order, the impedance

is also low.
At frequencies higher and lower than the
resonant frequency, the difference between
the reactances will be a definite quantity and
will add with the resistance to make the impedance higher and higher as the circuit is
tuned off the resonant frequency.
If Xc should be greater than XL, then the
term (XL
Xc) will give a negative number.
However, when the difference is squared the
product is always positive. This means that
the smaller reactance is subtracted from the

larger, regardless of whether it be capacitive
or inductive, and the difference squared.

RADIO

FREQUENCY

Figure 20
RESONANCE CURVE
Showing the increase in impedance at resonance for o parallel- resonant circuit, and similarly, the increase in current at resonance for
a series- rsonant circuit. The sharpness of
resonance is determined by the Q of the circuit,
as illustrated by a comparison between A,
B, and C.

Current and Voltage
in Series Resonant

Formulas for calculating
currents and voltages in
a series resonant circuit
are similar to those of

Circuits
Ohm's law.

=-Z
E

I

E =

IZ

The complete equations:
E

I
V' r= +

E =

1

(XL
+

(XL

Xc)2

Xc)'

Inspection of the above formulas will show
the following to apply to series resonant circuits: When the impedance is low, the current
will be high; conversely, when the impedance
is high, the current will be low.
Since it is known that the impedance will
be very low at the resonant frequency, it follows that the current will be a maximum at
this point. If a graph is plotted of the current
against the frequency either side of resonance,
the resultant curve becomes what is known as
a resonance curve. Such a curve is shown in
figure 20, the frequency being plotted against
current in the series resonant circuit.
Several factors will have an effect on the
shape of this resonance curve, of which re-

www.americanradiohistory.com

Circuit

HANDBOOK
sistance and L -to -C ratio are the important
considerations. The curves B and C in figure
20 show the effect of adding increasing values
of resistance to the circuit. It will be seen
that the peaks become less and less prominent
as the resistance is increased; thus, it can be
said that the selectivity of the circuit is
thereby decreased. Selectivity in this case
can be defined as the ability of a circuit to
discriminate against frequencies adjacent to
the resonant frequency.
Because the a.c. or r -f
voltage across a coil and
capacitor is proportional
to the reactance (for a
given current), the actual voltages across the
coil and across the capacitor may be many
times greater than the terminal voltage of the
circuit. At resonance, the voltage across the
coil (or the capacitor) is Q times the applied
voltage. Since the Q (or merit factor) of a
series circuit can be in the neighborhood of
100 or more, the voltage across the capacitor,
for example, may be high enough to cause
flashover, even though the applied voltage is
of a value considerably below that at which
the capacitor is rated.
Voltage Across Coil
and Capacitor in
Series Circuit

-Sharp-

extremely important
property of a capacitor or
an inductor is its factor of- merit, more generally called its Q. It is this
factor, Q, which primarily determines the
sharpness of resonance of a tuned circuit.
This factor can be expressed as the ratio of
the reactance to the resistance, as follows:
Circuit

Q

An

ness of Resonance

Q-

R

The actual resistance in a wire
or an inductor can be far greater
than the d-c value when the coil is used in a
radio -frequency circuit; this is because the
current does not travel through the entire
cross -section of the conductor, but has a tendency to travel closer and closer to the surface
of the wire as the frequency is increased. This
is known as the skin effect.
The actual current -carrying portion of the
wire is decreased, as a result of the skin
effect, so that the ratio of a -c to d -c resistance of the wire, called the resistance ratio,
is increased. The resistance ratio of wires to
be used at frequencies below about 500 kc.
may be materially reduced through the use of
Utz wire. Litz wire, of the type commonly used
to wind the coils of 455 -kc. i -f transformers,
may consist of 3 to 10 strands of insulated
wire, about No. 40 in size, with the individual
Skin

Effect

Examination of the equation
for determining Q might give
rise to the thought that even
though the resistance of an inductor increases
with frequency, the inductive reactance does
likewise, so that the Q might be a constant.
Actually, however, it works out in practice
that the Q of an inductor will reach a relatively broad maximum at some particular frequency.
Hence, coils normally are designed in such a
manner that the peak in their curve of Q with
frequency will occur at the normal operating
frequency of the coil in the circuit for which
it is designed.
The Q of a capacitor ordinarily is much
higher than that of the best coil. Therefore,
it usually is the merit of the coil that limits
the overall Q of the circuit.
At audio frequencies the core losses in an
iron -core inductor greatly reduce the Q from
the value that would be obtained simply by
dividing the reactance by the resistance. Obviously the core losses also represent circuit
resistance, just as though the loss occurred
in the wire itself.
Variation of Q
with Frequency

circuits, parallel resonance (more correctly termed anti resonance) is more frequently
encountered than series resonance; in fact, it
is the basic foundation of receiver and transmitter circuit operation. A circuit is shown in
figure 21.
Parallel

In radio

Resonance

"Tank" In this circuit, as contrasted with
a circuit for series resonance, L
(inductance) and C (capacitance)
are connected in parallel, yet the combination
can be considered to be in series with the
remainder of the circuit. This combination
of L and C, in conjunction with R, the resistance which is principally included in L, is
sometimes called a tank circuit because it effectively functions as a storage tank when incorporated in vacuum tube circuits.
Contrasted with series resonance, there are
two kinds of current which must be considered
in a parallel resonant circuit: (1) the line current, as read on the indicating meter M (2)
the circulating current which flows within the
parallel L -C -R portion of the circuit. See
figure 21.
At the resonant frequency, the line current
(as read on the meter M,) will drop to a very
low value although the circulating current in
the L -C circuit may be quite large. It is interesting to note that the parallel resonant circuit acts in a distinctly opposite manner to
that of a series resonant circuit, in which the
Circuit

where R = total resistance.

55

strands connected together only at the ends of
the coils.

The

2rrfL

Q

www.americanradiohistory.com

56

A!ternating Current Circuits

THE

RADIO

plifier circuit, the impedance curve must have
a sharp peak in order for the circuit to be
selective. If the curve is broad- topped in
shape, both the desired signal and the interfering signals at close proximity to resonance
will give nearly equal voltages on the grid of
the tube, and the circuit will then be nonselective; i.e., it will tune broadly.
Figure

21

PARALLEL- RESONANT CIRCUIT
The inductance L and capacitance C comprise
the reactive elements of the parallel -resonant
(anti -resonant) tank circuit, and the resistance
R indicates the sum of the r -f resistance of the
coil and capacitor, plus the resistance coupled
into the circuit from the external load. In most
coses the tuning capacitor has much lower r -f
resistance than the coil and can therefore be
ignored in comparison with the coil resistance
and the coupled -in resistance. The instrument
M1 indicates the "line current" which keeps
the circuit in a state of oscillation
current is the some as the fundamental component
of the plate current of a Class C amplifier which
might be feeding the tank circuit. The instrument M2 indicates the "tank current" which
is equal to the line current multiplied by the
operating Q of the tank circuit.

-this

current is at a maximum and the impedance is
minimum at resonance. It is for this reason
that in a parallel resonant circuit the principal
consideration is one of impedance rather than
current. It is also significant that the impedance curve for parallel circuits is very nearly
identical to that of the current curve for series
resonance. The impedance at resonance is
expressed as:

Z- (2trf

L)2

R

where Z = impedance in ohms,
L = inductance in henrys,
f = frequency in cycles,
R = resistance in ohms.
Or, impedance can be expressed as a function of Q as:

Z= 2nfLQ,
showing that the impedance of a circuit is
directly proportional to its effective Q at

resonance.
The curves illustrated in figure 20 can be
applied to parallel resonance. Reference to the
curve will show that the effect of adding resistance to the circuit will result in both a
broadening out and lowering of the peak of the
curve. Since the voltage of the circuit is
directly proportional to the impedance, and
since it is this voltage that is applied to the
grid of the vacuum tube in a detector or am-

highest
possible voltage can be
developed across a parallel resonant circuit, the impedance of this
circuit must be very high. The impedance will
be greater with conventional coils of limited
Q when the ratio of inductance-to-capacitance
is great, that is, when L is large as compared
with C. When the resistance of the circuit is
In order that the

Effect of L/C Ratio
in Parallel Circuits

very low. XL will equal XC at maximum impedance. There are innumerable ratios of L
and C that will have equal reactance, at a
given resonant frequency, exactly as in the
case in a series resonant circuit.
In practice, where a certain value of inductance is tuned by a variable capacitance
over a fairly wide range in frequency, the
L/C ratio will be small at the lowest frequency end and large at the high -frequency end.
The circuit, therefore, will have unequal gain
and selectivity at the two ends of the band of
frequencies which is being tuned. Increasing
the Q of the circuit (lowering the resistance)
will obviously increase both the selectivity
and gain.
Circulating Tank

The Q of a circuit has
definite bearing on
the
circulating tank
current at resonance. This tank current is
very nearly the value of the line current multiplied by the effective circuit Q. For example:
an r -f line current of 0.050 amperes, with a
circuit Q of 100, will give a circulating tank
current of approximately 5 amperes. From this
it can be seen that both the inductor and the
connecting wires in a circuit with a high Q
must be of very low resistance, particularly in
the case of high power transmitters, if heat
losses are to be held to a minimum.
Because the voltage across the tank at
resonance is determined by the Q, it is possible to develop very high peak voltages
across a high Q tank with but little line current.
Current at Resonance

Effect of Coupling
on Impedance

output circuit, the
Q of the parallel
coupling becomes
(tighter) coupling

www.americanradiohistory.com

a

If a parallel resonant circuit is coupled to another
circuit, such as an antenna
impedance and the effective
circuit is decreased as the
closer. The effect of closer
is the same as though an

Circuit Impedance

HANDBOOK

COUPLING
LOP

O vE N

4E011.14 COUPLING
MEDI U4 0

LOOSE COUPLING

HIGH 0

57

Z

t
Figure

EFFECT OF COUPLING

ON

actual resistance were added in series with
the parallel tank circuit. The resistance thus
coupled into the tank circuit can be considered as being reflected from the output or
load circuit to the driver circuit.
The behavior of coupled circuits depends
largely upon the amount of coupling, as shown
in figure 22. The coupled current in the secondary circuit is small, varying with frequency,
being maximum at the resonant frequency of
the circuit. As the coupling is increased
between the two circuits, the secondary resonance curve becomes broader and the resonant amplitude increases, until the reflected
resistance is equal to the primary resistance.
This point is called the critical coupling
point. With greater coupling, the secondary
resonance curve becomes broader and develops
double resonance humps, which become more
pronounced and farther apart in frequency as
the coupling between the two circuits is

increased.
Tank Circuit
Flywheel Effect

When the plate circuit of a
Class B or Class C operated

tube is connected to a parallel resonant circuit tuned to the same frequency as the exciting voltage for the amplifier, the plate current serves to maintain this
L/C circuit in a state of oscillation.
The plate current is supplied in short pulses

which do not begin to resemble a sine wave,
even though the grid may be excited by a sine wave voltage. These spurts of plate current
are converted into a sine wave in the plate
tank circuit by virtue of the "Q" or "flywheel
effect" of the tank.
If a tank did not have some resistance
losses, it would, when given a "kick" with a
single pulse, continue to oscillate indefinitely.
With a moderate amount of resistance or "fric-

tion" in

the circuit the tank

will still have

22

CIRCUIT IMPEDANCE AND

Q

inertia, and continue to oscillate with decreasing amplitude for a time after being given
a "kick." With such a circuit, almost pure

sine -wave voltage will be developed across
the tank circuit even though power is supplied
to the tank in short pulses or spurts, so long
as the spurts are evenly spaced with respect
to time and have a frequency that is the same
as the resonant frequency of the tank.
Another way to visualize the action of the
tank is to recall that a resonant tank with
moderate Q will discriminate strongly against
harmonics of the resonant frequency. The distorted plate current pulse in a Class C amplifier contains not only the fundamental frequency (that of the grid excitation voltage)
but also higher harmonics. As the tank offers
low impedance to the harmonics and high impedance to the fundamental (being resonant to
a sinethe latter), only the fundamental
appears across the tank circuit
wave voltage
in substantial magnitude.

-

-

Confusion sometimes exists as
to the relationship between the
unloaded and the loaded Q of the
tank circuit in the plate of an r -f power amplifier. In the normal case the loaded Q of the
tank circuit is determined by such factors as
the operating conditions of the amplifier, bandwidth of the signal to be emitted, permissible
level of harmonic radiation, and such factors.
The normal value of loaded Q for an r-f amplifier used for communications service is from
perhaps 6 to 20. The unloaded Q of the tank
circuit determines the efficiency of the output
circuit and is determined by the losses in the
tank coil, its leads and plugs and jacks if any,
and by the losses in the tank capacitor which
ordinarily are very low. The unloaded Q of a
good quality large diameter tank coil in the
high - frequency range may be as high as 500
Loaded and
Unloaded 0

www.americanradiohistory.com

THE

Alternating Current Circuits

58

to 800, and values greater than 300 are quite

-FUNDAMENTAL SINE WAVE(A

- FUNDAMENTAL
PLUS
3RD HARMONIC(C)

common.

-SQUARE WAVE

Tank Circuit

Since the unloaded Q of a tank
circuit is determined by the
minimum losses in the tank,
while the loaded Q is determined by useful
loading of the tank circuit from the external
load in addition to the internal losses in the
tank circuit, the relationship between the two
Q values determines the operating efficiency
of the tank circuit. Expressed in the form of
an equation, the loaded efficiency of a tank
Efficiency

3RD HARMONIC

Figure

circuit is:
Tank efficiency

=

1

_Qt

x 100

3 -3

23

FUNDAMENTAL PLUS 3RD HARMONIC

unloaded Q of the tank circuit
Qi = loaded Q of the tank circuit
As an example, if the unloaded Q of the
tank circuit for a class C r -f power amplifier
is 400, and the external load is coupled to the
tank circuit by an amount such that the loaded
Q is 20, the tank circuit efficiency will be:
eff. = (1 - 20/400) x 100, or (1 - 0.05) x 100,
or 95 per cent. Hence 5 per cent of the power
output of the Class C amplifier will be lost
as heat in the tank circuit and the remaining
95 per cent will be delivered to the load.
Qu

=

FUNDAMENTAL PLUS 3RD AND
5TH HARMONICS(E)
VI%

HARMONIC

(D)

Figure 24
THIRD HARMONIC WAVE PLUS
FIFTH HARMONIC
FUNDAMENTAL PLUS 3RD. 5TH,
AND 7TH HARMONICS
FUNDAMENTAL PLUS 3RD AND
STM HARMONICS
SQUARE WAVE

(G)

Nonsinusoidal Waves
and Transients

7TM HARMONIC

Pure sine waves,
basic wave shapes.
and complex shape
particularly square
and peaked waves.

(B)

COMPOSITE WAVE-FUNDAMENTAL
PLUS THIRD HARMONIC

Qu
where

RADIO

(F )

discussed previously, are
Waves of many different
are used in electronics,
waves, saw -tooth waves,

Any periodic wave (one that
repeats itself in definite
time intervals) is composed of sine waves of
different frequencies and amplitudes, added
together. The sine wave which has the same
frequency as the complex, periodic wave is
called the fundamental. The frequencies higher
than the fundamental are called harmonics,
and are always a whole number of times higher
than the fundamental. For example, the frequency twice as high as the fundamental is
called the second harmonic.

Wave Composition

The Square Wave

Figure

23 compares a square
wave with a sine wave (A)
of the same frequency. If another sine wave
(B) of smaller amplitude, but three times the
frequency of (A), called the third harmonic, is
added to (A), the resultant wave (C) more
nearly approaches the desired square wave.

Figure 25
RESULTANT WAVE, COMPOSED OF
FUNDAMENTAL, THIRD, FIFTH,
AND SEVENTH HARMONICS

This resultant curve (figure 24) is added to
fifth harmonic curve (D), and the sides of
the resulting curve (E)are steeper than before.
This new curve is shown in figure 25 after a
7th harmonic component has been added to it,
making the sides of the composite wave even
steeper. Addition of more higher odd harmonics
will bring the resultant wave nearer and nearer
a

to the desired square wave shape. The square
wave will be achieved if an infinite number of
odd harmonics are added to the original sine
wave.

www.americanradiohistory.com

Nonsinusoidal

HANDBOOK
FUND.PLUS 2ND ]RD, 4TH,
NAR MENICS
AND

FUND PLUS 210 HARM.
FUNDAMENTAL
2ND HARM.

Waves

59

FUNDAMENTAL PLUS
3RD HARMONIC
FUNDAMENTAL

PLUS 2ND 3RD,
iikkatIPUZTD"'
TM NARMOrMCS

]TM HARMONIC

3RD HARMONIC

-4111111
FVND.

HARMONICS

f YNO.

PLU12Np3RD

ATM\

STM AND STM NARMISN ICS

FUND, PLUS AND AND

PLUS END MARY.

3RD MARYON IC

FUND.

PLUS 2ND, 3RD ATM.

\AND STM MARMONIOS
T

TH HARMONIC

Z
FUND. PLUS 2ND, 3RD,
H XA RM ONI C4
S 2CS AN
A M

&IL"

FUND PLUS 2040, 3R0,4T14,
STM. TH, AND ?TM HARMS.
DL

S:w ÁNDU TMNthreCCaiTa

CATM

7TH HARMONIC

FUNDAMENTAL PLUS 3RD
AND STD HARMONICS

,FUNDAMENTAL PLUS 3RD HARM.
5TH HARMONIC
TOOTH WADE

SA

/

FUND. PLUS 2ND, 3RD4 ATM, STM, ATM,
AND 7TH HARMONICS

Figure 26
COMPOSITION OF A SAWTOOTH WAVE
FUNDAMENTAL PLUS 3RD, STN,
AND 7TH HARMONICS

FUNDAMENTAL PLUS 3RD
ANO STH HARMONIC

the same fashion, a
sawtooth wave is made up
of different sine waves (figure 26). The addition of all harmonics, odd and even, produces
the sawtooth wave form.
The Sawtooth Wave

In

7TH HARMONIC
/

Figure 27 shows the composition of a peaked wave.
Note how the addition of each sucessive harmonic makes the peak of the resultant higher
and the sides steeper.
The Pecked Wave

The three preceeding examples show how a complex
periodic wave is composed of a fundamental
wave and different harmonics. The shape of
the resultant wave depends upon the harmonics
that are added, their relative amplitudes, and
relative phase relationships. In general, the
steeper the sides of the waveform, the more
harmonics it contains.

Figure 27
COMPOSITION OF A PEAKED WAVE

Other Waveforms

If an a -c voltage is substituted for the d-c input voltage in the RC Transient circuits discussed in Chapter 2, the same principles may
be applied in the analysi* of the transient
behavior. An RC coupling circuit is designed
to have a long time constant with respect to
the lowest frequency it must pass. Such a
circuit is shown in figure 28. If a nonsinusoidal voltage is to be passed unchanged
through the coupling circuit, the time constant

AC Transient Circuits

must be long with respect to the period of the
lowest frequency contained in the voltage
wave.
An RC voltage divider that
is designed to distort the input waveform is known as a
differentiator or integrator, depending upon
the locations of the output taps. The output
from a differentiator is taken across the resistance, while the output from an integrator
is taken across the capacitor. Such circuits
will change the shape of any complex a-c
waveform that is impressed upon them. This
distortion is a function of the value of the
time constant of the circuit as compared to
the period of the waveform. Neither a differentiator nor an integrator can change the
RC

Differentiator

and

Integrator

www.americanradiohistory.com

60

Alternating Current Circuits
Cr

0.1 .Uf

100 v.
1000 C.P 5
R

THE

o.M

C'o.1 V

,00v
e.(PEAR)

OUTPUT
VOLTAGE

1000 C.P.S.

R'10R

cc.

RADIO

INTEGRATOR OUTPUT

1T

JJreR'DIPPERENTIATOROUTPUT

50000 USECONDS

R 1lC

PERIOD OP

e' 1000 USECONDS
+100V

Figure 28
R -C COUPLING CIRCUIT WITH
LONG TIME CONSTANT

100 v

et

,-11

INTEGRATOR
OUTPUT

I

E'100v.
(PEAR )

I

I

ófiE[Ñ1TÓR"N

eo

100

V

+25

V.

I

1

10001

-100v

eR

IATOR

DIP

OUTPUT

+30v
PR o

AO

T

OUTPUT Of

ATOR

o

(ec)

T

Figure

30

RC DIFFERENTIATOR AND
INTEGRATOR ACTION ON
Figure 29
R -C DIFFERENTIATOR AND
INTEGRATOR ACTION ON

A SQUARE WAVE

A SINE WAVE

Sawtooth Wave Input

shape of a pure sine wave, they will merely
shift the phase of the wave (figure 29). The
differentiator output is a sine wave leading
the input wave, and the integrator output is a
sine wave which lags the input wave. The sum
of the two outputs at any instant equals the

instantaneous input voltage.

If

a square wave voltage is
impressed on the circuit of
figure 30, a square wave voltage output may
be obtained across the integrating capacitor
if the time constant of the circuit allows the
capacitor to become fully charged. In this
particular case, the capacitor never fully
charges, and as a result the output of the
integrator has a smaller amplitude than the
input. The differentiator output has a maximum
value greater than the input amplitude, since
the voltage left on the capacitor from the
previous half wave will add to the input voltage. Such a circuit, when used as a differentiator, is often called a peaker. Peaks of
twice the input amplitude may be produced.

Square Wave Input

If

a back -to -back saw tooth voltage is applied
to an RC circuit having a time constant one sixth the period of the input voltage, the result is shown in figure 31. The capacitor
voltage will closely follow the input voltage,
if the time constant is short, and the integrator output closely resembles the input. The
amplitude is slightly reduced and there is a
slight phase lag. Since the voltage across the
capacitor is increasing at a constant rate, the
charging and discharging current is constant.
The output voltage of the differentiator, therefore, is constant during each half of the saw tooth input.

voltage waveforms
Various
other than those represented
here may be applied to short
RC circuits for the purpose of producing
across the resistor an output voltage with an
amplitude proportional to the rate of change of
the input signal. The shorter the RC time constant is made with respect to the period of the
input wave, the more nearly the voltage across
Miscellaneous
Inputs

www.americanradiohistory.com

Transformers

HANDBOOK

e=100

61

INTEGRATOR
OUTPUT (ec)

v.

(PEAR)
1000 C.P.S.

1 DIFFERENTIATOR
JOUTPUT

-

(e0)

+100

OUTPUT WAVEFORM
OF GENERATOR

-100

-ppt{ IO
uJ

e0

ÌA
ERÉITTÓRp
I(eR)

e0

OUTPUT OF
INTEGRATOR

(eE)

Figure 31
DIFFERENTIATOR AND
INTEGRATOR ACTION ON

R -C

A SAWTOOTH WAVE

Figure 32

the capacitor conforms to the input voltage.
Thus, the differentiator output becomes of
particular importance in very short RC circuits. Differentiator outputs for various types
of input waves are shown in figure 32.

The application of a square
wave input signal to audio
equipment, and the observation of the reproduced output signal on
an oscilloscope will provide a quick and accurate check of the overall operation of audio
equipment. Low -frequency and high- frequency
response, as well as transient response can be
examined easily. If the amplifier is deficient
in low- frequency response, the flat top of the
square wave will be canted, as in figure 33.
If the high- frequency response is inferior, the
rise time of the output wave will be retarded
(figure 34). An amplifier with a limited highand low- frequency response will turn the
square wave into the approximation of a saw tooth wave (figure 35).

Square Wave Test
for Audio Equipment

Transformers
are placed in such inductive
coils
When two
relation to each other that the lines of force
3-4

from one cut across the turns of the other
inducing a current, the combination can be
called a transformer. The name is derived from
the fact that energy is transformed from one
winding to another. The inductance in which

Dlfferentlator outputs of short r -c circuits for
various input voltage waveshapes. The output
voltage is proportional to the rote of change
of the input voltage.

the original flux is produced is called the
primary; the inductance which receives the
induced current is called the secondary. In a
radio receiver power transformer, for example,
the coil through which the 110 -volt a.c. passes
is the primary, and the coil from which a higher
or lower voltage than the a -c line potential is
obtained is the secondary.
Transformers can have either air or magnetic cores, depending upon the frequencies at
which they are to be operated. The reader
should thoroughly impress upon his mind the

fact that current can be transferred from one
circuit to another only if the primary current
is changing or alternating. From this it can be
seen that a power transformer cannot possibly
function as such when the primary is supplied
with non -pulsating d.c.
A power transformer usually has a magnetic
core which consists of laminations of iron,
built up into a square or rectangular form,
with a center opening or window. The secondary windings may be several in number, each
perhaps delivering a different voltage. The
secondary voltages will be proportional to the
turns ratio and the primary voltage.

www.americanradiohistory.com

Alternating Current Circuits

62

RADIO

THE

Figure 33
Amplifier deficient In low frequency response will distort square wave applied
to the input circuit, as
shown. A 60 -cycle square wave may he used.
A:
B:
C:
D:

Drop in gain at low frequencies
Lending phase shift at low frequencies
Logging phase shift at low frequencies
Accentuated low frequency gain

Types of
Transformers

Transformers are used in alteroaring- current circuits to transfer power at one voltage and impedance to another circuit at another voltage
and impedance. There are three main classifications of transformers: those made for use
in power-frequency circuits, those made
for
audio -frequency applications, and those made
for radio frequencies.
The Transformation
Ratio

In a perfect transformer all
the magnetic flux lines
produced by the primary
winding link every turn of the secondary winding. For such a transformer, the ratio of the
primary and secondary voltages is exactly
the same as the ratio of the number of turns
in the two windings:

Np

across the secondary
winding
In practice, the transformation ratio of a
transformer is somewhat less than the turns
ratio, since unity coupling does not exist
between the primary and secondary windings.
Ampere Turns (NI)

The current that flows in
the secondary winding as a
result of the induced voltage must produce a
flux which exactly equals the primary flux.
The magnetizing force of a coil is expressed
as the product of the number of turns in the
coil times the current flowing in it:

NpxIp =NsXIs,

or

Np

Ns

=

Is
IP

where Ip = primary current

Ep

Ns
Es
where Np = number of turns in the primary
winding
Ns = number of turns in the secondary
winding
Ep = voltage across the primary winding

Figure

Es = voltage

Is = secondary current
It can be seen from this expression that
when the voltage is stepped up, the current
is stepped down, and vice -versa.
Leakage Reactance

34

Since unity coupling does
not exist in a practical

Figure 35

Output waveshape of amplifier having deficiency
in high- frequency response. Tested with 10 -kc.
square wave.

Output waveshape of amplifier having limited
low -frequency and high- frequency response.
Tested with 1 -kc. square wave.

www.americanradiohistory.com

Electric Filters

HANDBOOK

63

1

STEP -UP

ZL
STEP -OOW N

INPUT

OUTPUT
VOLTAGE

VOLTAGE

Figure 36
IMPEDANCE -MATCHING TRANSFORMER
The reflected impedance Zp varies directly In

the secondary load IL, and
proportion to the square of the
primary-to- secondary turns ratio.

proportion

directly

to

In

transformer, part of the flux passing from the
primary circuit to the secondary circuit follows a magnetic circuit acted upon by the
primary only. The same is true of the secondary flux. These leakage fluxes cause leakage
reactance in the transformer, and tend to
cause the transformer to have poor voltage
regulation. To reduce such leakage reactance,
the primary and secondary windings should
be in close proximity to each other. The more
expensive transformers have interleaved windings to reduce inherent leakage reactance.
Impedance

In the ideal transformer, the

impedance of the secondary
load is reflected back into the
primary winding in the following relationship:

Transformation

Zp = N'Zs , or N = N/Zp/Zs
where Zp = reflected primary impedance
N = turns ratio of transformer
Zs = impedance of secondary load

Thus any specific load connected to the
secondary terminals of the transformer will
be transformed to a different specific value
appearing across the primary terminals of the
transformer. By the proper choice of turns
ratio, any reasonable value of secondary load
impedance may be "reflected" into the primary winding of the transformer to produce the
desired transformer primary impedance. The
phase angle of the primary "reflected" impedance will be the same as the phase angle
of the load impedance. A capacitive secondary load will be presented to the transformer
source as a capacity, a resistive load will
present a resistive "reflection" to the primary
source. Thus the primary source "sees" a
transformer load entirely dependent upon the
secondary load impedance and the turns ratio
of the transformer (figure 36).
The type of transformer in figure
37, when wound with heavy wire
over an iron core, is a common
device in primary power circuits for the purpose of increasing or decreasing the line volt-

Figure 37
THE AUTO -TRANSFORMER
auto- transformer
Schematic diagram of an
showing the method of connecting it to the line
and to the load. When only a small amount of

step up or step down Is required, the auto transformer may be much smaller physically
thon would be a transformer with o separate
Continuously variable
winding.
secondary
auto -transformers (Variar and Powerstat) are
widely used commercially.

age. In effect, it is merely a continuous winding with taps taken at various points along
the winding, the input voltage being applied
to the bottom and also to one tap on the winding. If the output is taken from this same
tap, the voltage ratio will be 1 -to -1; i.e., the
input voltage will be the same as the output
voltage. On the other hand, if the output tap
is moved down toward the common terminal,
there will be a step -down in the turns ratio
with a consequent step-down in voltage. The
initial setting of the middle input tap is chosen
so that the number of turns will have sufficient reactance to keep the no -load primary
current at a reasonably low value.

Electric Filters

3 -5

There are many applications where it is
desirable to pass a d -c component without
passing a superimposed a -c component, or to

ELEMENTARY FILTER SECTIONS
T- NET WONIt

L- SECTIONS

rs

T
Pi - NETWOR

The Auto

Transformer

Figure 38
Complex filters may be mode up from these basic

www.americanradiohistory.com

filter sections.

64

A l t e rn a t

n g

i

C ur r e n

LOW -PASS SHUNT -DERIVE

t

C

i

rc u

i

t

T H E

s

R A D

I

O

HIGH-PASS SERIES -DERI ED FILTER

FILTER

(SERIES -ARM RESONATED

(5J.UNT -ARM RESONATE

CI

2

2CI

2C1
C2

O
<
z

fQ

f2

fq

FREQUENCY

R.
L

FREQUENCY

LOAD RESISTANCE

R.

Ci
1

4M

x C

C2-

K

C2= MCK

LK=

f2 =

LOAD RESISTANCE

CI'

M LE

L2M=

,/I

-()2

CUT -OFF FREQUENCY.

Cx=

777_tF-

fk =FREQUENCY

OF

NIGH ATTENUATION

LK

14M>

-x

M-

=

Cs

7- M-I /ßa`2
(

fI=

I

coverage book.

Filter Operation A filter acts by virtue of its
property of offering very high
impedance to the undesired frequencies, while
offering but little impedance to the desired
frequencies. This will also apply to d.c. with
a superimposed a -c component, as d.c. can
be considered as an alternating current of zero
frequency so far as filter discussion goes.
Basic Filters

Filters are divided into four
classes, descriptive of the fre-

quency bands which they are designed to
transmit: high pass, low pass, band pass and
band elimination. Each of these classes of
filters is made up of elementary filter sections
called L sections which consist of a series
element (ZA) and a parallel element (ZR) as

477fIR

NIGH ATTENUATION

Figure 39
TYPICAL LOW -PASS AND HIGH -PASS FILTERS, ILLUSTRATING
DERIVATIONS

pass all frequencies above or below a certain
frequency while rejecting or attenuating all
others, or to pass only a certain band or bands
of frequencies while attenuating all others.
All of these things can be done by suitable
combinations of inductance, capacitance and
resistance. However, as whole books have
been devoted to nothing but electric filters, it
can be appreciated that it is possible only to
touch upon them superficially in a general

CS

Cur-OFF FREQUENCY. P, =FREQUENCY OF

SHUNT AND SERIES

illustrated in figure 38. A finite number of L
sections may be combined into basic filter
sections, called T networks or pi networks,
also shown in figure 38. Both the T and pi
networks may be divided in two to form halfsections.
Filter Sections

The most common filter section is one in which the two

impedances ZA and Zg are so related that
their arithmetical product is a constant: ZA x
Zg = K2 at all frequencies. This type of filter
section is called a constant-K section.
A section having a sharper cutoff frequency
than a constant -K section, but less attenuation at frequencies far removed from cutoff is
the M- derived section, so called because the
shunt or series element is resonated with a
reactance of the opposite sign. If the complementary reactance is added to the series arm,
the section is said to be shunt derived; if
added to the shunt arm, series derived. Each
impedance of the M- derived section is related
to a corresponding impedance in the constant K section by some factor which is a function
of the constant m. M, in turn, is a function of
the ratio between the cutoff frequency and
the frequency of infinite attenuation, and will

www.americanradiohistory.com

Filter Design

HANDBOOK
TT- SECTION FILTER DESIGN
M=0
CONSTANT K

R' LOAD

RESISTANCE

=CUT-OFF FREQUENCY

2

f.= FREQUENCY OFVERY
HIGH ATTENUATION

°---/-i
C

0

Ln- n R
f2

O

C2

ltf2

R

T

2

f2

RL

fICUT-OFFFREQUENCY
FREQUENCY OF VERY
HIGH ATTENUATION

2L2

LK=4

2L2--

21_2

4

/Vt

I

Cl. S1=
0.6

Lz= la
o.s

R

z

z
ó

0

/¡

='-( f12'oe
)

=

2L2

I

1

2L2 --

}Lt iLt

I
I

.`..1

t

I

2L2

0

SAME VALUES

ASM=0.6

1--k

.

1

fm
SAME CURVE AS M

<

\

c

0.6

z

ú

i

1

Lt=3.75L1t=t4M xLK

C/=Cn
L2=LK

H IGH PASS

Cn-

M=0_6

FREQUENCY

CI

LOAD RESISTANCE

o.6

VALUES AS M

--f2

<

.

}SAME

SAME CURVE AS

M

FREQUENCY

CK

o

f

fm

j

j
<

A

4M

Li
TiC2

ú

T

o o

z
0

0 6

0
t

i

C2=o.6Cn=Cn

z
0

aC

1

-O

Li

Ci

yI

t

Lt=0.6Ln=MLK
t-M2

Cz-Cn

.11-(--f-2-,. 2-

f

T

t------0
C2
T

t

Cn -

R.

OTO

Lt=Ln

LOW PASS

M

2

SECTIONSS
LFSEC

NG

-F-YO013y

I

65

f,

.<

r

FREQUENCY

/f

t

FREQUENCY

Figure 40
Through the use of the curves and equations which accompany the diagrams in the illustration above it is
possible to determine the correct values of inductance and capacitance for the usual types of pi- section

filters.

have some value between zero and one. As the
value of m approaches zero, the sharpness of
cutoff increases, but the less will be the
attenuation at several times cutoff frequency.
A value of 0.6 may be used for min most applications. The "notch" frequency is determined
by the resonant frequency of the tuned filter
element. The amount of attenuation obtained
at the "notch" when a derived section is used
is determined by the effective Q of the resonant arm (figure 39).
Filter Assembly

Constant -K sections and derived sections may be cas-

caded to obtain the combined characteristics
of sharp cutoff and good remote frequency
attenuation. Such a filter is known as a composite filter. The amount of attenuation will
depend upon the number of filter sections
used, and the shape of the transmission curve
depends upon the type of filter sections used.
All filters have some insertion loss. This
attenuation is usually uniform to all frequen-

cies within the pass band. The insertion loss
varies with the type of filter, the Q of the
components and the type of termination employed.

Electric wave filters have long
been used in some amateur siations in the audio channel to
reduce the transmission of unwanted high frequencies and hence to reduce the bandwidth
occupied by a radiophone signal. The effectiveness of a properly designed and properly
used filter circuit in reducing QRM and sideband splatter should not be underestimated.
In recent years, high frequency filters have
become commonplace in TVI reduction. High pass type filters are placed before the input
stage of television receivers to reject the
fundamental signal of low frequency transmitters. Low -pass filters are used in the output circuits of low frequency transmitters to
prevent harmonics of the transmitter from
being radiated in the television channels.
Electric Filter
Design

www.americanradiohistory.com

66

Alternating Current Circuits

The chart of figure 40 gives design data
and procedure on the pi- section type of filter.
M- derived sections with an M of 0.6 will be
found to be most satisfactory as the input
section (or half- section) of the usual filter
since the input impedance of such a section

is most constant over the pass band of the
filter section.
Simple filters may use either L, T, or n sections. Since the rr section is the more commonly used type figure 40 gives design data
and characteristics for this type of filter.

A PUSH -PULL 250 -TH AMPLIFIER WITH TVI SHIELD REMOVED
filters in power leads and antenna circuit reduces radiation of TVI- producing harmonics
of typical push -pull amplifier. Shielded enclosure completes harmonic reduction measures.

Use of harmonic

www.americanradiohistory.com

CHAPTE:R FOUR

Vacuum Tube Principles

electron tubes the cathode energy is applied
in the form of heat; electron emission from a
heated cathode is called tbermionic emission.
In another common type of electron tube, the
photoelectric cell, energy in the form of light
is applied to the cathode to cause photoelectric emission.

In the previous chapters we have seen the
manner in which an electric current flows
through a metallic conductor as a result of an
electron drift. This drift, which takes place
when there is a difference in potential between
the ends of the metallic conductor, is in addition to the normal random electron motion
between the molecules of the conductor.
The electron may be considered as a minute
negatively charged particle, having a mass of
9 x 10 -7° gram, and a charge of 1.59 x 10 -19
coulomb. Electrons are always identical,
regardless of the source from which they are

Thermionic Emission

4-1

of electrons from the
cathode of a thermionic electron
tube takes place when the cathode
of the tube is heated to a temperature sufficiently high that the free electrons in the
emitter have sufficient velocity to overcome
the restraining forces at the surface of the
material. These surface forces vary greatly
with different materials. Hence different types
of cathodes must be raised to different temperatures to obtain adequate quantities of electron emission. The several types of emitters
found in common types of transmitting and
receiving tubes will be described in the following paragraphs.
Electron
Emission

obtained.
An electric current can be caused to flow
through other media than a metallic conductor.
One such medium is an ionized solution, such
as the sulfuric acid electrolyte in a storage
battery. This type of current flow is called
electrolytic conduction. Further, it was shown
at about the turn of the century that an electric current can be carried by a stream of free
electrons in an evacuated chamber. The flow
of a current in such a manner is said to take
place by electronic conduction. The study of
electron tubes (also called vacuum tubes, or
valves) is actually the study of the control and
use of electronic currents within an evacuated
or partially evacuated chamber.
Since the current flow in an electron tube
takes place in an evacuated chamber, there
must be located within the enclosure both a
source of electrons and a collector for the
electrons which have been emitted. The electron source is called the cathode, and the
electron collector is usually called the anode.
Some external source of energy must be applied to the cathode in order to impart sufficient velocity to the electrons within the
cathode material to enable them to overcome
the surface forces and thus escape into the
surrounding medium. In the usual types of

Emission

Cathode Types The emitters or cathodes as
used in present -day thermionic electron tubes may be classified into
two groups: the directly- heated or filament
type and the indirectly -heated or heater - cathode
type. Directly- heated emitters may be further

subdivided into three important groups, all
of which are commonly used in modern vacuum
tubes. These classifications are: the puretungsten filament, the thoriated- tungsten
filament, and the oxide-coated filament.
The Pure Tung-

sten Filament

67

www.americanradiohistory.com

Pure tungsten wire was used
as the filament in nearly all
the earlier transmitting and

68

Vacuum

Tube

THE

Principles

RADIO

Figure
ELECTRON TUBE TYPES
The new General E l e c t r i c ceramic triode (68Y4) is shown alongside a conventional miniature tube (6265) and an octal -based receiving tube (25L6). The ceramic
tube is designed for rugged service and features extremely low lead inductance.
1

receiving tubes. However, the thermionic efficiency of tungsten wire as an emitter (the
number of milliamperes emission per watt of
filament heating power) is quite low, the filaments become fragile after use, their life is
rather short, and they are susceptible to burnout at any time. Pure tungsten filaments must
be run at bright white heat (about 2500° Kelvin). For these reasons, tungsten filaments
have been replaced in all applications where
another type of filament could be used. They
are, however, still universally employed in
large water-cooled tubes and in certain large,
high -power air- cooled triodes where another
filament type would be unsuitable. Tungsten
filaments are the most satisfactory for high power, high -voltage tubes where the emitter
is subjected to positive ion bombardment
caused by the residual gas content of the
tubes. Tungsten is not adversely affected by
such bombardment.
the course of experiments made upon tungsten
emitters, it was found that
filaments made from tungsten having a small
amount of thoria (thorium oxide) as an impurity had much greater emission than those
made from the pure metal. Subsequent development has resulted in the highly efficient carburized thoriated- tungsten filament as used in
virtually all medium -power transmitting tubes
today.
Thoriated-tungsten emitters consist of a
tungsten wire containing from 1% to 2% thoria.
The activation process varies between different manufacturers of vacuum tubes, but
it is essentially as follows: (1) the tube is
evacuated; (2) the filament is burned for a
short period at about 2800° Kelvin to clean
the surface and reduce some of the thoria
within the filament to metallic thorium; (3)
The ThoriatedTungsten Filament

In

the filament is burned for a longer period at
about 2100° Kelvin to form a layer of thorium on the surface of the tungsten; (4) the
temperature is reduced to about 1600° Kelvin
and some pure hydrocarbon gas is admitted
to form a layer of tungsten carbide on the
surface of the tungsten. This layer of tungsten
carbide reduces the rate of thorium evaporation from the surface at the normal operating
temperature of the filament and thus increases
the operating life of the vacuum tube. Thorium evaporation from the surface is a natural
consequence of the operation of the thoriatedtungsten filament. The carburized layer on the
tungsten wire plays another role in acting as
a reducing agent to produce new thorium from
the thoria to replace that lost by evaporation. This new thorium continually diffuses to
the surface during the normal operation of
the filament. The last process, (5), in the
activation of a thoriated tungsten filament consists of re- evacuating the envelope and then
burning or ageing the new filament for a considerable period of time at the normal operating temperature of approximately 1900°K.
One thing to remember about any type of
filament, particularly the thoriated type, is
that the emitter deteriorates practically as
fast when "standing by" (no plate current) as
it does with any normal amount of emission
load. Also, a thoriated filament may be either
temporarily or permanently damaged by a
heavy overload which may strip the surface
layer of thorium from the filament.

Thoriated- tungsten filaments (and only thoriatedtungsten filaments) which
have lost emission as
a result of insufficient filament voltage, a
severe temporary overload, a less severe extended overload, or even normal operation
Reactivating
Thoriated- Tungsten
Filaments

www.americanradiohistory.com

Types of Emitters

HANDBOOK

Figure

69

2

V -H -F and U -H -F TUBE TYPES

The tube to the left In this photograph is a 955 "acorn" triode. The 6F4 acorn triode is very similar in
appearance to the 955 but has two leads brought out each for the grid and for the plate connection. The
second tubs Is a 446A "lighthouse" triode. The 2C40, 2C43, and 2C44 ore more recent examples of the
same type tube and are
tially the same in external appearance. The third tube from the left is o
2C39 "oilcan" tube. This tube type is essentially the inverse of the lighthouse variety since the cathode
and heater connections come out the small end and the plats is the large finned radiator on the large end.
The use of the finned plate radiator makes the oilcan tube capable of approximately 10 times as much
plate dissipation as the lighthouse type. The tube to the right is the 4X 150A beam tetrads. This tube, a
comparatively recent release, is capable of somewhat greater power output than any of the other tube
types shown, and is rated for full output at 500 Mc. and of reduced output at frequencies greater than
1000 Mc.

may quite frequently be

reactivated to their
original characteristics by a process similar
to that of the original activation. However,
only filaments which have not approached too
close to the end of their useful life may be
successfully reactivated.
The actual process of reactivation is relatively simple. The tube which has gone
"flat" is placed in a socket to which only the
two filament wires have been connected. The
filament is then "flashed" for about 20 to 40
seconds at about 1% times normal rated voltage. The filament will become extremely bright
during this time and, if there is still some
thoria left in the tungsten and if the tube did
not originally fail as a result of an air leak,
some of this thoria will be reduced to metallic
thorium. The filament is then burned at 15 to
25 per cent overvoltage for from 30 minutes to
3 to 4 hours to bring this new thorium to the
surface.
The tube should then be tested to see if it
shows signs of renewed life. If it does, but is
still weak, the burning process should be continued at about 10 to 15 per cent overvoltage
for a few more hours. This should bring it
back almost to normal. If the tube checks still
very low after the first attempt at reactivation,
the complete process can be repeated as a
last effort.
The Oxide.
Coated Filament

The most efficient of all
modern filaments
is the
oxide-coated type which con-

sists of a mixture of barium and strontium
oxides coated upon a nickel alloy wire or
strip. This type of filament operates at a dull red to orange -red temperature (1050' to 1170°
K) at which temperature it will emit large

quantities of electrons. The oxide -coated
filament is somewhat more efficient than the
thoriated- tungsten type in small sizes and it
is considerably less expensive to manufacture.
For this reason all receiving tubes and quite
a number of the low -powered transmitting
tubes use the oxide - coated filament. Another
advantage of the oxide -coated emitter is its
the average tube can be
extremely long life

-

expected to run from 3000 to 5000 hours, and
when loaded very lightly, tubes of this type
have been known to give 50,000 hours of life
before their characteristics changed to any
great extent.

Oxide filaments are unsatisfactory for use
at high continuous plate voltages because: (1)
their activity is seriously impaired by the
high temperature necessary to de -gas the high voltage tubes and, (2) the positive ion bombardment which takes place even in the best
evacuated high -voltage tube causes destruction of the oxide layer on the surface of the

filament.
Oxide -coated emitters have been found capable of emitting an enormously large current
pulse with a high applied voltage for a very
short period of time without damage. This
characteristic has proved to be of great value

www.americanradiohistory.com

70

Vacuum

Tube

Figure 3
CUTAWAY DRAWING OF

A

THE

Principles

6C4 TRIODE

RADIO

age from 2 to 117 volts, although 6.3 is the
most common value. The heater is operated
at quite a high temperature so that the cathode
itself usually may be brought to operating
temperature in a matter of 15 to 30 seconds.
Heat -coupling between the heater and the
cathode is mainly by radiation, although there
is some thermal conduction through the insulating coating on the heater wire, as this
coating is also in contact with the cathode
thimble.
Indirectly heated cathodes are employed in
all a-c operated tubes which are designed to
operate at a low level either for r-f or a -f use.
However, some receiver power tubes use
heater cathodes (6L6, 6V6, 6F6, and 6K6 -GT)
as do some of the low-power transmitter tubes
(802, 807, 815, 3E29, 2E26, 5763, etc.). Heater
cathodes are employed almost exclusively
when a number of tubes are to be operated in
series as in an a.c. -d.c. receiver. A heater
cathode is often called a uni- potential cathode
because there is no voltage drop along its
length as there is in the directly- heated or
filament cathode.

in radar work. For example, the relatively
small cathode in a microwave magnetron may
be called upon to deliver 25 to 50 amperes at
an applied voltage of perhaps 25,000 volts for
a period in the order of one microsecond.
After this large current pulse has been passed,
plate voltage normally will be removed for
1000 microseconds or more so that the cathode
surface may be restored in time for the next
pulse of current. If the cathode were to be
subjected to a continuous current drain of this
magnitude, it would be destroyed in an exceedingly short period of time.
The activation of oxide- coated filaments
also varies with tube manufacturers but consists essentially in heating the wire which has
been coated with a mixture of barium and
strontium carbonates to a temperature of about
1500° Kelvin for a time and then applying a
potential of 100 to 200 volts through a protective resistor to limit the emission current.
This process reduces the carbonates to oxides
thermally, cleans the filament surface of
foreign materials, and activates the cathode

special bombardment cathode is employed in many of
the new high powered television transmitting klystrons(Eimac 3K 20,000
LA). The cathode takes the form of a tantalum
diode, heated to operating temperature by the
bombardment of electrons from a directly
heated filament. The cathode operates at a
positive potential of 2000 volts with respect
to the filament, and a d-c bombardment current of 0.66 amperes flows between filament
and cathode. The filament is designed to
operate under space -charge limited conditions.
Cathode temperature is varied by changing the
bombardment potential between the filament
and the cathode.

The heater type cathode was developed as a result of the requirement for a type of emitter
which could be operated from alternating current and yet would not introduce a-c ripple
modulation even when used in low -level stages.
It consists essentially of a small nickel -alloy
cylinder with a coating of strontium and barium oxides on its surface similar to the coating used on the oxide- coated filament. Inside
the cylinder is an insulated heater element
consisting usually of a double spiral of tungsten wire. The heater may operate on any volt-

The emission of electrons from
a heated cathode is quite similar to the evaporation of molecules from the surface of a liquid. The molecules which leave the surface are those having
sufficient kinetic (heat) energy to overcome
the forces at the surface of the liquid. As the
temperature of the liquid is raised, the average velocity of the molecules is increased,
and a greater number of molecules will acquire
sufficient energy to be evaporated. The evaporation of electrons from the surface of a thermionic emitter is similarly a function of average electron velocity, and hence is a function
of the temperature of the emitter.
Electron emission per unit area of emitting
surface is a function of the temperature T
in degrees Kelvin, the work function of the
emitting surface b (which is a measure of the

surface.
Reactivation of oxide- coated filaments is
not possible since there is always more than
sufficient reduction of the oxides and diffusion
of the metals to the surface of the filament to
meet the emission needs of the cathode.
The Heater
Cathode

The Bombardment
Cathode

The Emission

Equation

www.americanradiohistory.com

A

Thermionic Emission

HANDBOOK

71

600

TYPE

6CB6

6W4-GT

Er' 63 VOLTS

/

PLATE

600

35LP?RS314R

w

ÌSCREEN

W

4910

__-, frT.

¢

400

çArHOOE HEATER
HEATER
L

-

-

-

--

i

,

F

ww

200

á
Figure

4

CUT -AWAY DRAWING OF A 6CB6 PENTODE

surface forces of the material and hence of
the energy required of the electron before it
may escape), and of the constant A which
also varies with the emitting surface. The relationship between emission current in amperes per square centimeter, 1, and the above
quantities can be expressed as:
= AT'c"b'T
Secondary The bombarding of most metals
and a few insulators by electrons
Emission
will result in the emission of other
1

electrons

by a

process called secondary emis-

The secondary electrons are literally
knocked from the surface layers of the bombarded material by the primary electrons which
strike the material. The number of secondary
electrons emitted per primary electron varies
from a very small percentage to as high as
5 to 10 secondary electrons per primary.
The phenomena of secondary emission is
undesirable for most thermionic electron tubes.
However, the process is used to advantage in
certain types of electron tubes such as the
image orthicon (TV camera tube) and the
electron -multiplier type of photo -electric cell.
In types of electron tubes which make use of
secondary emission, such as the type 931
photo cell, the secondary- electron -emitting
surfaces are specially treated to provide a
high ratio of secondary to primary electrons.
Thus a high degree of current amplification in
the electron -multiplier section of the tube is
sion.

obtained.
As a cathode is heated so that
it begins to emit, those electrons which have been discharged into the surrounding space form a
negatively charged cloud in the immediate
vicinity of the cathode. This cloud of electrons
around the cathode is called the space charge.
The electrons comprising the charge are conThe Space
Charge Effect

tinuously

changing,

since those electrons

making up the original charge fall back into

20

10

30

ao

so

D.C. PLATE VOLTS

Figure

5

AVERAGE PLATE CHARACTERISTICS
OF A POWER DIODE

the cathode and are replaced by others emitted
by it.
4 -2

The Diode

If a cathode capable of being heated either
indirectly or directly is placed in an evacuated
envelope along with a plate, such a two element vacuum tube is called a diode. The
diode is the simplest of all vacuum tubes and
is the fundamental type from which all the
others are derived.

the cathode within a
diode is heated, it will be
found that a few of the electrons leaving the cathode will leave with sufficient velocity to reach the plate. If the plate
is electrically connected back to the cathode,
the electrons which have had sufficient veloc=
ity to arrive at the plate will flow back to the
cathode through the external circuit. This
small amount of initial plate current is an
effect found in all two -element vacuum tubes.
If a battery or other source of d -c voltage
is placed in the external circuit between the
plate and cathode so that it places a positive
potential on the plate, the flow of current from
the cathode to plate will be increased. This is
due to the strong attraction offered by the positively charged plate for any negatively charged
particles (figure 5).
Characteristics
of the Diode

When

At moderate values of
plate voltage the current flow from cathode
to anode is limited by the space charge of
electrons around the cathode. Increased values

Space- Charge Limited

Current

www.americanradiohistory.com

Vacuum

72

Principles

Tube

THE

RADIO

DE COATED

ED TUNGSTEN

TUNGSTEN FILAMENT
POINT OF MAXIMUM SPACE CHARGE -LIMITED EMISSION

Figure
ACTION

PLATE VOLTAGE

Figure

+

6

MAXIMUM SPACE -CHARGE -LIMITED
EMISSION FOR DIFFERENT

TYPES OF EMITTERS

of plate voltage will tend to neutralize a
greater portion of the cathode space charge
and hence will cause a greater current to flow.
Under these conditions, with plate current
limited by the cathode space charge, the plate
current is not linear with plate voltage. In
fact it may be stated in general that the plate current flow in electron tubes does not obey
Ohm's Law. Rather, plate current increases as
the three -halves power of the plate voltage.
The relationship between plate voltage, E,
and plate current, 1, can be expressed as:
/

=K

F3!2

where K is a constant determined by the
geometry of the element structure within the
electron tube.
As plate voltage is raised to
the potential where the cathode space charge is neutralized, all the electrons that the cathode is capable of emitting are being attracted to the
plate. The electron tube is said then to have
reached saturation plate current. Further increase in plate voltage will cause only a
relatively small increase in plate current. The
initial point of plate current saturation is
sometimes called the point of Maximum Space Charge- Limited Emission (MSCLE).
The degree of flattening in the plate -voltage
plate- current curve after the MSCLE point will
vary with different types of cathodes. This effect is shown in figure 6. The flattening is
quite sharp with a pure tungsten emitter. With
thoriated tungsten the flattening is smoothed
somewhat, while with an oxide- coated cathode
the flattening is quite gradual. The gradual
saturation in emission with an oxide- coated
emitter is generally considered to result from
Plate Current
Saturation

7

OF

THE GRID IN A TRIODE
(A) shows the triode tube with cutoff bias on
the grid. Note that all the electrons emitted
by the cathode remain inside the grid mesh.
(B) shows the same tube with an intermediate
value of bias on the grid. Note the medium
value of plate current and the fact that there
is a reserve of electrons remaining within the
grid mesh. (C) shows the operation with a
relatively small amount of bias which with
certain tube types will allow substantially all
the electrons emitted by the cathode to reach
the plate. Emission is said to be saturated in
this case. In a majority of tube types a high
value of positive grid voltage is required before plate - current saturation takes place.

a lowering of the surface work function by the
field at the cathode resulting from the plate

potential.

Electron Energy
Dissipation

The current flowing in the
plate- cathode space of a conducting electron tube represents the energy required to accelerate electrons from the zero potential of the cathode
space charge to the potential of the anode.
Then, when these accelerated electrons strike
the anode, the energy associated with their
velocity is immediately released to the anode
structure. In normal electron tubes this energy
release appears as heating of the plate or
anode structure.
4-3

The Triode

If an element consisting of a mesh or spiral
of wire is inserted concentric with the plate
and between the plate and the cathode, such
an element will be able to control by electrostatic action the cathode -to -plate current of
the tube. The new element is called a grid, and
a vacuum tube containing a cathode, grid, and
plate is commonly called a triode.
Action of

If this new element through which
the electrons must pass in their
course from cathode to plate is made
negative with respect to the cathode, the negathe Grid

www.americanradiohistory.com

HANDBOOK

mom

TYPE

N
4

MI

6J5

Er=6.5VOLTS

A

litiliI
MM1I
1111111
italar

!
A
/!!.

.111!

..01

;

!

/I
AMÍGlI
o

100

test.

pp.

200

300

IM
400

Current Flow
in a Triode

500

PLATE VOLTS (EP)

Figure 8
NEGATIVE -GRID CHARACTERISTICS(Ip
VS. Ep CURVES) OF A

TYPICAL

TRIODE
Average plate characteristics of this type
are most commonly used in determining the
Class A operating characteristics of a
triode amplifier stage.

cive charge on this grid will effectively repel
the negatively charged electrons (like charges
repel; unlike charges attract) back into the

space charge surrounding the cathode Hence,
the number of electrons which are able to pass
through the grid mesh and reach the plate will
be reduced, and the plate current will be reduced accordingly. If the charge on the grid
is made sufficiently negative, all the electrons
leaving the cathode will be repelled back to
it and the plate current will be reduced to zero.
Any d-c voltage placed upon a grid is called
a bias (especially so when speaking of a control grid). The smallest negative voltage which
will cause cutoff of plate current at a particular plate voltage is called the value of cutoff
bias (figure 7).

Amplification The amount of plate current in a
Factor
triode is a result of the net field
at the cathode from interaction
between the field caused by the grid bias and
that caused by the plate voltage. Hence, both

grid bias and plate voltage affect the plate
current. In all normal tubes a small change in
grid bias has a considerably greater effect
than a similar change in plate voltage. The
ratio between the change in grid bias and the
change in plate current which will cause the
same small change in plate current is called
the amplification /actor or
of the electron
tube. Expressed as an equation:

AE

- AE,

73

with i, constant (A represents a small increment).
The µ can be determined experimentally by
making a small change in grid bias, thus
slightly changing the plate current. The plate
current is then returned to the original value
by making a change in the plate voltage. The
ratio of the change in plate voltage to the
change in grid voltage is the µ of the tube
under the operating conditions chosen for the

16

s!

Characteristics

Triode

In a diode it was shown that

the electrostatic field at the
cathode was proportional to
the plate potential, Ep, and that the total
cathode current was proportional to the three halves power of the plate voltage. Similiarly,
in a triode it can be shown that the field at
the cathode space charge is proportional to
the equivalent voltage (Eg + Ep /ft), where
the amplification factor, µ, actually represents
the relative effectiveness of grid potential and
plate potential in producing a field at the
cathode.
It would then be expected that the cathode
current in a triode would be proportional to
the three -halves power of (E5 + Ep /µ). The
cathode current of a triode can be represented
with fair accuracy by the expression:

Cathode current

= K

(E5

E

3/2

+-=-)
)

where K is a constant determined by element
geometry within the triode.

Plate Resistance The plate resistance of a
vacuum tube is the ratio of a
change in plate voltage to the change in plate
current which the change in plate voltage
produces. To be accurate, the changes should
be very small with respect to the operating
values. Expressed as an equation:
Rp =

AE P

p

E,=

constant, A = small

increment
The plate resistance can also be determined
by the experiment mentioned above. By noting
the change in plate current as it occurs when
the plate voltage is changed (grid voltage
held constant), and by dividing the latter by
the former, the plate resistance can be determined. Plate resistance is expressed in Ohms.
The mutual conductance,
also referred to as trans conductance, is the ratio of a change in the
plate current to the change in grid voltage
which brought about the plate current change,
the plate voltage being held constant. Expressed as an equation:
Transconductance

www.americanradiohistory.com

Tube

Vacuum

74

Principles

RADIO

THE

430

400
330
300

2
110

_
W
6
6

seo

W

130

C

100

-20
o

-0 -e -10

10

10

b

/0 40
10
GRID VOLTAGE (E9)

70

e0

10

100

characteristics of this type are most
commonly used in determining the pulse -signal
operating characteristics of a triode amplifier
stage. Note the large emission capability of
the oxidecoated heater cathode in tubes of the
type of the 6J5.
9

Al

constant,

small
increment
The transconductance is also numerically
equal to the amplification factor divided by
the plate resistance. Gm =
E

=

A =

AEg

Transconductance is most commonly expressed in microreciprocal -ohms or micro mhos. However, since transconductance expresses change in plate current as a function
of a change in grid voltage, a tube is often
said to have a transconductance of so many
milliamperes- per-volt. If the transconductance
in milliamperes -per-volt is multiplied by 1000
it will then be expressed in micromhos. Thus
the transconductance of a 6A3 could be called
either 5.25 ma.!volt or 5250 micromhos.

0

1-5

(Es)

Figure 10

This type of graphical
for Class C amplifier
operating characteristic
is a straight line when

Plate

1

-10
-5
GRID VOLTS

CONSTANT CURRENT (Ep VS. E9)
CHARACTERISTICS OF A
TYPICAL TRIODE TUBE

Figure 9
POSITIVE -GRID CHARACTERISTICS
(Ip VS. E9) OF A TYPICAL TRIODE

Gm =

-11

representation is used
calculations since the
of a Class C amplifier
drawn upon a constant

current- graph.

passing through the plate circuit of the tube
for various values of plate -load resistance and
plate - supply voltage. Figure 11 illustrates a
triode tube with a resistive plate load, and a
supply voltage of 300 volts. The voltage at
the plate of the tube (ep) may be expressed
as:

ep = Ep

-(i

x RL)

where Ep is the plate supply voltage, ip is the
plate current, and RL is the load resistance in
ohms.
Assuming various values of ip flowing in
the circuit, controlled by the internal resistance of the tube, (a function of the grid bias)
values of plate voltage may be plotted as
shown for each value of plate current (ir). The
line connecting these points is called the
load line for the particular value of plate -load
resistance used. The slope of the load line is
equal to the ratio of the lengths of the vertical
and horizontal projections of any segment of
the load line. For this example it is:

The operating character istics of a triode tube
may be summarized in
three sets of curves: The Ip vs. Ep curve
(figure 8), the Ip vs. Eg curve (figure 9) and
the Ep vs. E curve (figure 10). The plate
resistance (Rj of the tube may be observed
from the Ip vs. Ep curve, the transconductance
(Gm) may be observed from the Ip vs. Eg curve,
and the amplification factor (pt) may be determined from the Ep vs. Eg curve.

The slope of the load line is equal to
-1/11L. At point A on the load line, the voltage across the tube is zero. This would be
true for a perfect tube with zero internal voltage drop, or if the tube is short -circuited from
cathode to plate. Point B on the load line

load line is a graphical
representation of the voltage
on the plate of a vacuum tube, and the current

corresponds to the cutoff point of the tube,
where no plate current is flowing. The operating range of the tube lies between these
two extremes. For additional information re-

Characteristic Curves

of a Triode Tube

The Load Line

A

Slope

=

-

www.americanradiohistory.com

.01
100

-

.02
200

-

.0001

-

1

10,000

Triode Load Line

HANDBOOK
IP(YA)i En
S

10

IS

20
25
30

EP=300v

I

300
250
200
I

75

RL=en

SO

100
50
0

Figure 12
TRIODE TUBE CONNECTED FOR DETERMINATION OF PLATE CIRCUIT LOAD
LINE, AND OPERATING PARAMETERS
OF THE CIRCUIT

300

EP

Figure

11

voltage drop across the plate load resistor,
RL. The plate voltage on the tube is therefore
300 volts. If, on the other hand, the tube is
considered to be a short circuit, maximum
possible plate current flows and the full 300
volt drop appears across RL. The plate voltage is zero, and the plate current is 300/8,000,
or 37.5 milliamperes. These two extreme conditions define the load line on the I, vs. Ep
characteristic curve, figure 13.
For this application the grid of the tube is
returned to a steady biasing voltage of -4
volts. The steady or quiescent operation of the
tube is determined by the intersection of the
load line with the -4 volt curve at point Q.
By projection from point Q through the plate

The static load line for a typical triode
tube with a plate load resistance of 10,000
ohms.

garding dynamic load lines, the reader is
referred to the Radiotron Designer's Handbook,
4th edition, distributed by Radio Corporation
of America.
Application of Tube
Characteristics

As an example of the ap-

plication of tube characteristics, the constants
of the triode amplifier circuit shown in figure
12

volts, and the plate load is 8,000 ohms.

If the tube is considered to be an open circuit
no plate current will flow, and there is no

may be considered. The plate supply is

40
37.5

3s
0J.

30

,
--i , ..
LOGO LINO

4000n

iAJ

1

2

Figure

=

EG

fL

INSTANTANEOUS

SWING

13

i
2
2

APPLICATION OF Ip VS. Ep
CHARACTERISTICS OF
VACUUM TUBE

IP11145-410.2

dFt

,t'

-- --

14N.
o

leo

x

2ee

s

300

400

PLATE VOLTS (E.)

N

www.americanradiohistory.com

VOLT Pt.AT[ SWING

`

Vacuum

76

EG

-4r

Tube

\

Principles

. D.C.

THE

RADIO

BIAS LEVEL (EC)

TFigure 15
SCHEMATIC REPRESENTATION

+ 18.23

OF
STEADY STATE //
PLATE CURRENT\i)1

INTERELECTRODE
CAPACITANCE

+is
comes apparent. A voltage variation of 8 volts
(peak -to -peak) on the grid produces a variation
of 84 volts at the plate.

T-

Polarity Inversion

STEADY STATE (EP)
PLATE VOLTAGE

EP

Figure

14

POLARITY REVERSAL BETWEEN GRID
AND PLATE VOLTAGES

current axis it is found that the value of plate
current with no signal applied to the grid is
12.75 milliamperes. By projection from point
Q through the plate voltage axis it is found
that the quiescent plate voltage is 198 volts.
This leaves a drop of 102 volts across RL
which is borne out by the relation 0.01275 x
8,000 = 102 volts.
An alternating voltage of 4 volts maximum
swing about the normal bias value of -4 volts
is applied now to the grid of the triode amplifier. This signal swings the grid in a positive
direction to 0 volts, and in a negative direction
to -8 volts, and establishes the operating
region of the tube along the load line between
points A and B. Thus the maxima and minima
of the plate voltage and plate current are
established. By projection from points A and
B through the plate current axis the maximum
instantaneous plate current is found to be
18.25 milliamperes and the minimum is 7.5
milliamperes. By projections from points A and
B through the plate voltage axis the minimum
instantaneous plate voltage swing is found to
be 154 volts and the maximum is 240 volts.
By this graphical application of the IP vs.
Ep characteristic of the 6SN7 triode the operation of the circuit illustrated in figure 12 be-

signal voltage applied to the grid has its
maximum positive instantaneous value the
plate current is also maximum. Reference to
figure 12 shows that this maximum plate current flows through the plate load resistor RL,
producing a maximum voltage drop across it.
The lower end of RL is connected to the plate
supply, and is therefore held at a constant
potential of 300 volts. With maximum voltage
drop across the load resistor, the upper end of
RL is at a minimum instantaneous voltage.
The plate of the tube is connected to this end
of RL and is therefore at the same minimum
instantaneous potential.
This polarity reversal between instantaneous
grid and plate voltages is further clarified by
a consideration of Kirchhoff's law as it applies to series resistance. The sum of the IR
drops around the plate circuit must at all times
equal the supply voltage of 300 volts. Thus
when the instantaneous voltage drop across
RL is maximum, the voltage drop across the
tube is minimum, and their sum must equal
300 volts. The variations of grid voltage,plate
current and plate voltage about their steady
state values is illustrated in figure 14.
When the

Capacitance always exists between any two pieces of metal
separated by a dielectric. The
exact amount of capacitance depends upon the
size of the metal pieces, the dielectric between them, and the type of dielectric. The
electrodes of a vacuum tube have a similar
characteristic known as the interelectrode
capacitance, illustrated in figure 15. These
direct capacities in a triode are: grid-tocathode capacitance, grid -to-plate capacitance,
and plate -to- cathode capacitance. The interelectrode capacitance, though very small, has
a coupling effect, and often can cause unbalance in a particular circuit. At very high
Interelectrode
Capacitance

www.americanradiohistory.com

Tetrodes

HANDBOOK

Pentodes

and

TYPE 24-A

77

G=-3

esc =so v.

u
cr
u

6

TYPE 6SK7
esc= too v.
esu =ov.
`

O.

4

4

J

200

300

S00

100

VOLTS (Eel

Figure

TYPICAL

16

200

300

VOLTS

(E)

Figure

TETRODE
CHARACTERISTIC CURVES
Ip VS. Ep

frequencies (v-h -f), interelectrode capacities
become very objectionable and prevent the use
of conventional tubes at these frequencies.
Special v -h -f tubes must be used which are
characterized by very small electrodes and
close internal spacing of the elements of the
tube.

400

500

17

TYPICAL IP VS. EP PENTODE
CHARACTERISTIC CURVES
the electrons pass through it and on to the
plate. Due also to the screen, the plate current is largely independent of plate voltage,
thus making for high amplification. When the
screen voltage is held at a constant value, it
is possible to make large changes in plate
voltage without appreciably affecting the plate

current, (figure 16).
4 -4

Tetrode or Screen Grid Tubes

Many desirable characteristics can be obtained in a vacuum tube by the use of more
than one grid. The most common multi -element
tube is the tetrode (four electrodes). Other
tubes containing as many as eight electrodes
are available for special applications.

The quest for a simple and easily
usable method of eliminating the
effects of the grid-to -plate capacitance of the
triode led to the development of the screen grid tube or tetrode. When another grid is
added between the grid and plate of a vacuum
tube the tube is called a tetrode, and because
the new grid is called a screen, as a result of
its screening or shielding action, the tube is
often called a screen -grid tube. The interposed screen grid acts as an electrostatic
shield between the grid and plate, with the
consequence that the grid-to -plate capacitance
is reduced. Although the screen grid is maintained at a positive voltage with respect to
the cathode of the tube, it is maintained at
ground potential with respect to r.f. by means
of a by-pass capacitor of very low reactance
at the frequency of operation.
In addition to the shielding effect, the
screen grid serves another very useful purpose.
Since the screen is maintained at a positive
potential, it serves to increase or accelerate
the flow of electrons to the plate. There being
large openings in the screen mesh, most of
The Tetrode

When the electrons from the cathode approach the plate with sufficient velocity, they
dislodge electrons upon striking the plate.
This effect of bombarding the plate with high
velocity electrons, with the consequent dislodgement of other electrons from the plate,
gives rise to the condition of secondary emission which has been discussed in a previous
paragraph. This effect can cause no particular
difficulty in a triode because the secondary

electrons so emitted are eventually attracted
back to the plate. In the screen -grid tube, however, the screen is close to the plate and is
maintained at a positive potential. Thus, the
screen will attract these electrons which have
been knocked from the plate, particularly when
the plate voltage falls to a lower value than
the screen voltage, with the result that the
plate current is lowered and the amplification
is decreased.
In the application of tetrodes, it is necessary to operate the plate at a high voltage in
relation to the screen in order to overcome
these effects of secondary emission.
The undesirable effects of secondary emission from the plate
can be greatly reduced if yet another element
is added between the screen and plate. This
additional element Is called a suppressor, and
tubes in which it is used are called pentodes.
The suppressor grid is sometimes connected
to the cathode within the tube; sometimes it is
brought out to a connecting pin on the tube
base, but in any case it is established negaThe Pentode

www.americanradiohistory.com

Vacuum

78

Tube

Principles

THE

RADIO

GRiD
-

C

HODE

.L-

Cÿ

REMOTE CUT -OFF
GRID

SHARP CUT -OFF
GRID

-

Figure 18
REMOTE CUTOFF GRID STRUCTURE

tive with respect to the minimum plate voltage. The secondary electrons that would travel

to the screen if there were no suppressor are
diverted back to the plate. The plate current
is, therefore, not reduced and the amplification possibilities are increased (figure 17).
Pentodes for audio applications are designed so that the suppressor increases the
limits to which the plate voltage may swing;
therefore the consequent power output and
gain can be very great. Pentodes for radio frequency service function in such a manner
that the suppressor allows high voltage gain,
at the same time permitting fairly high gain
at low plate voltage. This holds true even if
the plate voltage is the same or slightly lower
than the screen voltage.
Remote cutoff tubes (variable
are screen grid tubes in
which the control grid structure has been physically modified so as to
cause the plate current of the tube to drop off
gradually, rather than to have a well defined
cutoff point (figure 18). A non -uniform control
grid structure is used, so that the amplification factor is different for different parts of the
Remote Cutoff

mu)

Tubes

control grid.
Remote cutoff tubes are used in circuits
where it is desired to control the amplification
by varying the control grid bias. The characteristic curve of an ordinary screen grid tube
has considerable curvature near the plate current cutoff point, while the curve of a remote
cutoff tube is much more linear (figure 19).
The remote cutoff tube minimizes cross-

talk

interference that would otherwise be
produced. Examples of remote cutoff tubes
are: 6BD6, 6K7, 6SG7 and 6SK7.
beam power tube makes use
of another method for suppressing
secondary emission. In this tube
there are four electrodes: a cathode, a grid, a
screen, and a plate, so spaced and placed that
secondary emission from the plate is suppressed without actual power loss. Because
Beam Power

Tubes

A

GRID VOLTS

Figure 19
A REMOTE CUTOFF
GRID STRUCTURE

ACTION OF

of the manner in which the electrodes are
spaced, the electrons which travel to the
plate are slowed down when the plate voltage
is low, almost to zero velocity in a certain
region between screen and plate. For this
reason the electrons form a stationary cloud,
or space charge. The effect of this space
charge is to repel secondary electrons emitted
from the plate and thus cause them to return
to the plate. In this way, secondary emission
is suppressed.
Another feature of the beam power tube is
the low current drawn by the screen. The
screen and the grid are spiral wires wound so
that each turn in the screen is shaded from
the cathode by a grid turn. This alignment of
the screen and the grid causes the electrons
to travel in sheets between the turns of the
screen so that very few of them strike the
screen itself. This formation of the electron
stream into sheets or beams increases the
charge density in the screen -plate region and
assists in the creation of the space charge in
this region.
Because of the effective suppressor action
provided by the space charge, and because of
the low current drawn by the screen, the beam
power tube has the advantages of high power
output, high power -sensitivity, and high efficiency. The 6L6 is such a beam power tube,
designed for use in the power amplifier stages
of receivers and spec -h amplifiers or modulators. Larger tubes employing the beam -power
principle are being made by various manufacturers for use in the radio -frequency stages
of transmitters. These tubes feature extremely
high power- sensitivity (a very small amount
of driving power is required for a large output), good plate efficiency, and low grid -toplate capacitance. Examples of these tubes
are 813, 4 -250A, 4X150A, etc.
The grid- screen mu factor (Ass)
is analogous to the amplification
factor in a triode, except that
the screen of a pentode or tetrode is subGrid -Screen
Mu

Factor

www.americanradiohistory.com

HANDBOOK

Mixer and Converter

stituted for the plate of a triode. µ5g denotes
the ratio of a change in grid voltage to a

change in screen voltage, each of which will
produce the same change in screen current.
Expressed as an equation:
AEss
flag =

AEs

Ise = constant, A = small

increment

The grid- screen mu factor is important in
determining the operating bias of a tetrode
or pentode tube. The relationship between control -grid potential and screen potential determines the plate current of the tube as well as
the screen current since the plate current is
essentially independent of the plate voltage
in tubes of this type. In other words, when
the tube is operated at cutoff bias as determined by the screen voltage and the grid screen mu factor (determined in the same way
as with a triode, by dividing the operating
voltage by the mu factor) the plate current
will be substantially at cutoff, as will be the
screen current. The grid- screen mu factor is
numerically equal to the amplification factor
of the same tetrode or pentode tube when
it is triode connected.
The following equation is the
expression for total cathode cur rent in a triode tube. The expression for the total cathode
current of a tetrode and a pentode tube is the
same, except that the screen -grid voltage and
the grid- screen it-factor are used in place of
the plate voltage and it of the triode.
Current Flow
in Tetrodes
and Pentodes

Cathode current = K

/
1

E

Es +

3/2

$g )
Ilse

Cathode current, of course, is the sum of the
screen and plate current, plus control grid current in the event that the control grid is positive with respect to the cathode. It will be

noted that total cathode current is independent
of plate voltage in a tetrode or pentode. Also,
in the usual tetrode or pentode the plate current is substantially independent of plate
voltage over the usual operating range- which
means simply that the effective plate resistance of such tubes is relatively high. However, when the plate voltage falls below the
normal operating range, the plate current
falls sharply, while the screen current rises to
such a value that the total cathode current
remains substantially constant. Hence, the
screen grid in a tetrode or pentode will almost
invariably be damaged by excessive dissipation if the plate voltage is removed while the
screen voltage is still being applied from a
low -impedance source.

Tubes

79

The current equations show how
the total cathode current in
triodes, tetrodes, and pentodes
is a function of the potentials applied to the
various electrodes. If only one electrode is
positive with respect to the cathode (such as
would be the case in a triode acting as a
class A amplifier) all the cathode current goes
to the plate. But when both screen and plate
are positive in a tetrode or pentode, the cathode current divides between the two elements.
Hence the screen current is taken from the
total cathode current, while the balance goes
to the plate. Further, if the control grid in a
tetrode or pentode is operated at a positive
potential the total cathode current is divided
between all three elements which have a positive potential. In a tube which is receiving a
large excitation voltage, it may be said that
the control grid robs electrons from the output
electrode during the period that the grid is
positive, making it always necessary to limit
the peak -positive excursion of the control
grid.

The Effect of
Grid Current

In general it may be stated
that the amplification factor
of tetrode and pentode tubes
is a coefficient which is not
of much use to the designer. In fact the amplification factor is seldom given on the design
data sheets of such tubes. Its value is usually
very high, due to the relatively high plate
resistance of such tubes, but bears little
relationship to the stage gain which actually
will be obtained with such tubes.
On the other hand, the grid-plate transconductance is the most important coefficient of
pentode and tetrode tubes. Gain per stage can
be computed directly when the Gm is known.
The grid -plate transconductance of a tetrode
or pentode tube can be calculated through use
of the 'expression:
Alp
Coefficients of
Tetrodes and
Pentodes

Gm

=

AE e

with E5s and Es constant.
The plate resistance of such tubes is of
less importance than in the case of triodes,
though it is often of value in determining the
amount of damping a tube will exert upon the
impedance in its plate circuit. Plate resistance is calculated from:
AEp
R
v

with Es and Esg constant.
4 -5

The

Mixer and Converter Tubes

superheterodyne receiver always in-

www.americanradiohistory.com

80

Vacuum Tube

Principles

THE

RADIO

OSCILLATOR GRID

- PLATE

r_FSCREEN

CAT NODE

GRID

METAL SPELL

FILAMENT

`

SUPPRESSOR

AND SHELL

Figure

SIGNAL GRID

Figure 20

LEAD INDUCTANCE

GRID STRUCTURE OF 6SA7

The degenerative action of cathode lead inductance tends to reduce the effective grid-tocathode voltage with respect to the voltage
available across the input tuned circuit. Cathode lead inductance also introduces undesirable coupling between the input and the out-

CONVERTER TUBE

eludes at least one stage for changing the
frequency of the incoming signal to the fixed
frequency of the main intermediate amplifier
in the receiver. This frequency changing
process is accomplished by selecting the
beat -note difference frequency between a
locally generated oscillation and the incoming
signal frequency. If the oscillator signal is
supplied by a separate tube, the frequency
changing tube is called a mixer. Alternatively,
the oscillation may be generated by additional
elements within the frequency changer tube.
In this case the frequency changer is commonly called a converter tube.
Conversion
Conductance

The conversion conductance(Ge)
is a coefficient of interest in the
case of mixer or converter tubes,
or of conventional triodes, tetrodes, or pentodes operating as frequency changers. The
conversion conductance is the ratio of a
change in the signal -grid voltage at the input
frequency to a change in the output current at
the converted frequency. Hence Gc in a mixer
is essentially the same as transconductance
in an amplifier, with the exception that the
input signal and the output current are on different frequencies. The value of G, in conventional mixer tubes is from 300 to 1000
micromhos. The value of G, in an amplifier
tube operated as a mixer is approximately 0.3
the Gm of the tube operated as an amplifier.
The voltage gain of a mixer stage is equal to
GCZL where ZL is the impedance of the plate
load into which the mixer tube operates.
The simplest mixer tube is
the diode. The noise figure,
or figure of merit, for a mixer of this type is
not as good as that obtained with other more
complex mixers; however, the diode is useful
as a mixer in u -h -f and v -h -f equipment where
low interelectrode capacities are vital to circuit operation. Since the diode impedance is
The Diode Mixer

21

SHOWING THE EFFECT OF CATHODE

put circuits.

low, the local oscillator must furnish considerable power to the diode mixer. A good
diode mixer has an overall gain of about 0.5.
A triode mixer has better
gain and a better noise figure
than the diode mixer. At low frequencies, the
gain and noise figure of a triode mixer closely
approaches those figures obtained when the
tube is used as an amplifier. In the u -h -f and
v -h -f range, the efficiency of the triode mixer
deteriorates rapidly. The optimum local oscillator voltage for a triode mixer is about 0.7 as
large as the cutoff bias of the triode. Very
little local oscillator power is required by a
triode mixer.
The Triode Mixer

Pentode Mixers and
Converter Tubes

The most common multi grid converter tube for
broadcast or shortwave
use is the penta grid converter, typified by
the 6SA7, 6SB7 -Y and 6BA7 tubes (figure 20).
Operation of these converter tubes and pentode
mixers will be covered in the Receiver Fundamentals Chapter.

4 -6

Electron Tubes at Very
High Frequencies

As the frequency of operation of the usual
type of electron tube is increased above about
20 Mc., certain assumptions which are valid
for operation at lower frequencies must be reexamined. First, we find that lead inductances
from the socket connections to the actual
elements within the envelope no longer are
negligible. Second, we find that electron

www.americanradiohistory.com

HANDBOOK

The

transit time no longer may be ignored; an
appreciable fraction of a cycle of input signal
may be required for an electron to leave the
cathode space charge, pass through the grid
wires, and travel through the space between
grid and plate.
The effect of lead inductance is two -fold. First, as
shown in figure 21, the
combination of grid -lead inductance, gridcathode capacitance, and cathode lead inductance tends to reduce the effective grid- cathode
signal voltage for a constant voltage at the
tube terminals as the frequency is increased.
Second, cathode lead inductance tends to
introduce undesired coupling between the
various elements within the tube.
Tubes especially designed for v -h -f and
u -h -f use have had their lead inductances
minimized. The usual procedures for reducing
lead inductance are: (1) using heavy lead
conductors or several leads in parallel (examples are the 6SH7 and 6AK5), (2) scaling
down the tube in all dimensions to reduce
both lead inductances and interelectrode
capacitances (examples are the 6AK5, 6F4,
and other acorn and miniature tubes), and (3)
the use of very low inductance extensions of
the elements themselves as external connections (examples are lighthouse tubes such as
the 2C40, oilcan tubes such as the 2C29, and
many types of v -h -f transmitting tubes).
Effects of
Lead Inductance

Effect of
Transit Time

When

erated

an electron tube is opat a frequency high

enough that electron transit
time between cathode and plate is an appreciable fraction of a cycle at the input frequency, several undesirable effects take place.
First, the grid takes power from the input
signal even though the grid is negative at all
times. This comes about since the grid will
have changed its potential during the time
required for an electron to pass from cathode
to plate. Due to interaction, and a resulting
phase difference between the field associated
with the grid and that associated with a moving electron, the grid presents a resistance to
an input signal in addition to its normal
"cold" capacitance. Further, as a result of
this action, plate current no longer is in phase
with grid voltage.
An amplifier stage operating at a frequency
high enough that transit time is appreciable:
(a) Is difficult to excite as a result of grid
loss from the equivalent input grid resistance,
(b) Is capable of less output since transconductance is reduced and plate current is
not in phase with grid voltage.
The effects of transit time increase with the
square of the operating frequency, and they

Klystron

81

increase rapidly as frequency is increased
above the value where they become just appreciable. These effects may be reduced by
scaling down tube dimensions; a procedure
which also reduces lead inductance. Further,
transit-time effects may be reduced by the
obvious procedure of increasing electrode potentials so that electron velocity will be increased. However, due to the law of electron motion in an electric field, transit time is
increased only as the square root of the ratio
of operating potential increase; therefore this
expedient is of limited value due to other
limitations upon operating voltages of small
electron tubes.
4 -7

Special Microwave
Electron Tubes

Due primarily to the limitation imposed by
transit time, conventional negative -grid electron tubes are capable of affording worthwhile
amplification and power output only up to a
definite upper frequency. This upper frequency
limit varies from perhaps 100 Mc. for conventional tube types to about 4000 Mc. for
specialized types such as the lighthouse tube.

Above the limiting frequency, the conventional
negative -grid tube no longer is practicable and
recourse must be taken to totally different
types of electron tubes in which electron
transit time is not a limitation to operation.
Three of the most important of such microwave
tube types are the klystron, the magnetron, and
the travelling wave tube.

The klystron is a type
of electron tube in which
electron transit time is used to advantage,
Such tubes comprise, as shown in figure 22,
a cathode, a focussing electrode, a resonator
connected to a pair of grids which afford
velocity modulation of the electron beam
(called the "buncher "), a drift space, and
another resonator connected to a pair of grids
(called the "catcher "). A collector for the
expended electrons may be included at the
end of the tube, or the catcher may also perform the function of electron collection.
The tube operates in the following manner:
The cathode emits a stream of electrons which
is focussed into a beam by the focussing
electrode. The stream passes through the
buncher where it is acted upon by any field
existing between the two grids of the buncher
cavity. When the potential between the two
grids is zero, the stream passes through without change in velocity. But when the potential
between the two grids of the buncher is increasingly positive in the direction of electron
The Power Klystron

www.americanradiohistory.com

82

Vacuum

TWO- CAVITY
A conventional
with a feedback
two cavities so

Tube

Principles

Figure 22
KLYSTRON OSCILLATOR
two -cavity klystron is shown
loop connected between the
that the tube may be used as
an

oscillator.

motion, the velocity of the electrons in the
beam is increased. Conversely, when the field
becomes increasingly negative in the direction
of the beam (corresponding to the other half
cycle of the exciting voltage from that which
produced electron acceleration) the velocity
of the electrons in the beam is decreased.
When the velocity- modulated electron beam
reaches the drift space, where there is no field,
those electrons which have been sped up on
one half -cycle overtake those immediately
ahead which were slowed down on the other
half- cycle. In this way, the beam electrons become bunched together. As the bunched groups
pass through the two grids of the catcher
cavity, they impart pulses of energy to these
grids. The catcher grid -space is charged to
different voltage levels by the passing electron
bunches, and a corresponding oscillating field
is set up in the catcher cavity. The catcher is
designed to resonate at the frequency of the
velocity- modulated beam, or at a harmonic of
this frequency.
In the klystron amplifier, energy delivered
by the buncher to the catcher grids is greater
than that applied to the buncher cavity by the
input signal. In the klystron oscillator a feedback loop connects the two cavities. Coupling
to either buncher or catcher is provided by
small Loops which enter the cavities by way of

concentric lines.
The klystron is an electron -coupled device.
When used as an oscillator, its output voltage
is rich in harmonics. Klystron oscillators of
various types afford power outputs ranging
from less than I watt to many thousand watts.
Operating efficiency varies between 5 and 30
per cent. Frequency may be shifted to some
extent by varying the beam voltage. Tuning is

THE

RADIO

Figure 23
REFLEX KLYSTRON OSCILLATOR
A conventional reflex klystron oscillator of
the type commonly used as o local oscillator
in superheterodyne receivers operating above
about 2000 Mc. is shown above. Frequency
modulation of the output frequency of the oscillator, or o-f-c operation in a receiver, may be
obtained by varying the negative voltage on the

repeller electrode.

carried on mechanically in some klystrons by
altering (by means of knob settings) the shape
of the resonant cavity.
two -cavity klystron
as described in the preceding paragraphs is primarily used as a transmitting device since quite reasonable amounts
of power are made available in its output circuit. However, for applications where a much
smaller amount of power is required-power
levels in the milliwatt range for low -power
transmitters, receiver local oscillators, etc.,
another type of klystron having only a single
cavity is more frequently used.
The theory of operation of the single- cavity
klystron is essentially the same as the multi cavity type with the exception that the velocity- modulated electron beam, after having left
the " buncher" cavity is reflected back into
the area of the buncher again by a repeller
electrode as illustrated in figure 23. The
potentials on the various electrodes are adjusted to the value such that proper bunching
of the electron beam will take place just as a
particular portion of the velocity -modulated
beam reenters the area of the resonant cavity.
Since this type of klystron has only one circuit
it can be used only as an oscillator and not as
an amplifier. Effective modulation of the frequency of a single- cavity klystron for FM
work can be obtained by modulating the repeller electrode voltage.
The Reflex Klystron

www.americanradiohistory.com

The

-

HANDBOOK

Magnetron

The
PLATE

83

MAGNET COIL

1

ANODE

ANODE

FIL
GRID
TERMINAL

ANODE
TERMINAL
II`

CATHODE

/

IL

ANODE
GLASS

`PLATE 2

SEAL

GLASS ENVELOPE

O

ANODE

OR'D

HEATER

FILAMENT
VOLTAGE

EYELET

SEAL

LEAD
TERMINAL

EYELET

TURULATiON

lower

Figure 24
CUTAWAY VIEW OF
WESTERN ELECTRIC 416- B/6280
VHF PLANAR TRIODE TUBE
The 416 -B, designed by the Bell
Telephone Laboratories is intended
for amplifier or frequency multiplier
service in the 4000 me region. Employing grid wires having a diameter
equal to fifteen wavelengths of light,
416 -B

has a transconductance of

50,000.
Spacing between grid and
cathode is .0005', to reduce transit
time effects. Entire tube is gold plated.

The Magnetron

magnetron is an
oscillator tube normally
The

PLATE
VOLTAGE

Figure 25
SIMPLE MAGNETRON OSCILLATOR
An external tank circuit is used with this type
of magnetron oscillator for operation in the

GLASS

the

FILAMENT

s -h -f

em-

ployed where very high values of peak power
or moderate amounts of average power are
required in the range from perhaps 700 Mc.
to 30,000 Mc. Special magnetrons were developed for wartime use in radar equipments
which had peak power capabilities of several
million watts (megawatts) output at frequencies in the vicinity of 3000 Mc. The normal
duty cycle of operation of these radar equipments was approximately 1 /10 of one per
cent (the tube operated about 1 /1000 of the
time and rested for the balance of the operating period) so that the average power output
of these magnetrons was in the vicinity of
1000 watts.

u -h

-f ronge.

In its simplest form the magnetron tube is a
filament -type diode with two half-cylindrical
plates or anodes situated coaxially with respect to the filament. The construction is
illustrated in figure 25A. The anodes of the
magnetron are connected to a resonant circuit
as illustrated on figure 25B. The tube is surrounded by an electromagnet coil which, in
turn, is connected to a low -voltage d -c energizing source through a rheostat R for controlling the strength of the magnetic field. The
field coil is oriented so that the lines of
magnetic force it sets up are parallel to the
axis of the electrodes.
Under the influence of the strong magnetic
field, electrons leaving the filament are deflected from their normal paths and move in
circular orbits within the anode cylinder. This
effect results in a negative resistance which
sustains oscillations. The oscillation frequency is very nearly the value determined by
L and C. In other magnetron circuits, the frequency may be governed by the electron rotation, no external tuned circuits being employed. Wavelengths of less than 1 centimeter have been produced with such circuits.
More complex magnetron tubes employ no
external tuned circuit, but utilize instead one
or more resonant cavities which are integral
with the anode structure. Figure 26 shows a
magnetron of this type having a multi -cellular

www.americanradiohistory.com

84

Vacuum

Tube

Principles
-

RADIO

THE
WAVE GUIDE

CATNODE LEAOE

WAVE GU IDE
OUTPUT

INPUT

ELECTRON BEAM
MAGNETRON

PE MANE NT

CTNODE

MAGNET

ANODE ESSASS

IIII

rT TING Outryt

NODE BLOC

iii1!,;,'+

Figure 26
MODERN MULTI- CAVITY MAGNETRON
Illustrated is an external -anode strapped magnetron of the type commonly used in radar equipment for the 10 -cm. range. A permanent magnet
of the general type used with such a magnetron
Is shown in the right -hand portion of the drawing,
with the magnetron in place between the pole
pieces of the magnet.

anode of eight cavities. It will be noted, also,
that alternate cavities (which would operate at
the same polarity when the tube is oscillating)
are strapped together. Strapping was found to
improve the efficiency and stability of high power radar magnetrons. In most radar applications of magnetron oscillators a powerful
permanent magnet of controlled characteristics
is employed to supply the magnetic field
rather than the use of an electromagnet.
The Travelling
Wave Tube

Travelling Wave Tube
(figure 27) consists of a helix
located within an evacuated
envelope. Input and output terminations are
affixed to each end of the helix. An electron
beam passes through the helix and interacts
with a wave travelling along the helix to produce broad band amplification at microwave
frequencies.
When the input signal is applied to the gun
end of the helix, it travels along the helix wire
at approximately the speed of light. However,
the signal velocity measured along the axis
of the helix is considerably lower. The electrons emitted by the cathode gun pass axially
through the helix to the collector, located at
the output end of the helix. The average velocity of the electrons depends upon the potential
of the collector with respect to the cathode.
When the average velocity of the electrons is
greater than the velocity of the helix wave,
the electrons become crowded together in the
various regions of retarded field, where they
impart energy to the helix wave. A power gain
of 100 or more may be produced by this tube.
4 -8

The

The Cathode -Ray Tube

The Cathode -Ray Tube

cathode -ray

The

is

a

tube

special type of

ANODE

COLLECTOR

Figure 27
THE TRAVELLING WAVE TUBE
Operation of this tube is the result of inter.
action between the electron beam and wave
travelling along the helix.

electron tube which permits the visual observation of electrical signals. It may be incorporated into an oscilloscope for use as a test
instrument or it may be the display device for
radar equipment or a television receiver.

cathode -ray tube always includes an electron gun for producing a stream of electrons, a
grid for controlling the intensity of the electron beam, and a luminescent screen for converting the impinging electron beam into visible light. Such a tube always operates in conjunction with either a built -in or an external
means for focussing the electron stream into a
narrow beam, and a means for deflecting the
electron beam in accordance with an electrical
signal.
The main electrical difference between
types of cathode -ray tubes lies in the means
employed for focussing and deflecting the
electron beam. The beam may be focussed
and/or deflected either electrostatically or
magnetically, since a stream of electrons can
be acted upon either by an electrostatic or a
magnetic field. In an electrostatic field the
electron beam tends to be deflected toward the
positive termination of the field (figure 28).
In a magnetic field the stream tends to be
deflected at right angles to the field. Further,
an electron beam tends to be deflected so that
it is normal (perpendicular) to the equipotential
lines of an electrostatic field- and it tends to
be deflected so that it is parallel to the lines
of force in a magnetic field.
Large cathode -ray tubes used as kinescopes
in television receivers usually are both focused
and deflected magnetically. On the other hand,
the medium -size CR tubes used in oscilloscopes and small television receivers usually
are both focused and deflected electrostatically. But CR tubes for special applications
may be focused magnetically and deflected
electrostatically or vice versa.
There are advantages and disadvantages to
Operation of
the CRT

A

www.americanradiohistory.com

HANDBOOK

The
- NOR

ACCELERATING ANODE IN)

SASE
NEATE

C

Lr.

LEG T

DE IFI

I

LONTAL DEFLECTION
ICI

OA

r1

-ADUADAG

SECONDARY

COATING

-1

è.

CLCCTIINNEEAu

RUORESCENT

CONTROL ACCELERATI
ANODE (A)
GRID IG
CATNODE (10

SCREEN

ZATNODE

-VERTICAL DEFLECTION
PLATES IS)

Figure 28

TYPICAL ELECTROSTATIC
CATHODE -RAY TUBE

both types of focussing and deflection. However, it may be stated that electrostatic deflection is much better than magnetic deflection
when high -frequency waves are to be displayed
on the screen; hence the almost universal use
of this type of deflection for oscillographic
work. But when a tube is operated at a high
value of accelerating potential so as to obtain
a bright display on the face of the tube as for
television or radar work, the use of magnetic
deflection becomes desirable since it is relatively easier to deflect a high -velocity electron
than electrostatically.
beam magnetically
However, an ion trap is required with magnetic deflection since the heavy negative ions
emitted by the cathode are not materially deflected by the magnetic field and hence would
burn an "ion spot" in the center of the luminescent screen. With electrostatic deflection
the heavy ions are deflected equally as well
as the electrons in the beam so that an ion
spot is not formed.
Construction of
The construction of a typical
Electrostatic CRT electrostatic- focus, electrostatic- deflection cathode-ray
tube is illustrated in the pictorial diagram of
figure 28. The indirectly heated cathode K releases free electrons when heated by the
enclosed filament. The cathode is surrounded
by a cylinder G, which has a small hole in its
front for the passage of the electron stream.
Although this element is not a wire mesh as
is the usual grid, it is known by the same
name because its action is similar: it controls
the electron stream when its negative potential

is varied.
Next in order, is found the first accelerating
anode, H, which resembles another disk or
cylinder with a small hole in its center. This
electrode is run at a high or moderately high
positive voltage, to accelerate the electrons
towards the far end of the tube.
The focussing electrode, F, is a sleeve
which usually contains two small disks, each
with a small hole.
After leaving the focussing electrode, the
electrons pass through another accelerating

Cathode

Ray

Tube

85

anode, A, which is operated at a high positive
potential. In some tubes this electrode is operatcd at a higher potential than the first accelerating electrode, H, while in other tubes both
accelerating electrodes are operated at the
same potential.

The electrodes which have been described
this point constitute the electron gun,
which produces the free electrons and focusses
them into a slender, concentrated, rapidly traveling stream for projecting onto the viewing screen.
up to

Electrostatic
Deflection

To make the tube useful, means
must be provided for deflecting
the electron beam along two axes
at right angles to each other. The more corn mon tubes employ electrostatic deflection
plates, one pair to exert a force on the beam
in the vertical plane and one pair to exert a
force in the horizontal plane. These plates
are designated as B and C in figure 28.
Standard oscilloscope practice with small
cathode -ray tubes calls for connecting one of
the B plates and one of the C plates together
and to the high voltage accelerating anode.
With the newer three -inch tubes and with five inch tubes and larger, all four deflecting plates
are commonly used for deflection. The positive
high voltage is grounded, instead of the negative as is common practice in amplifiers, etc.,
in order to permit operation of the deflecting
plates at a d-c potential at or near ground.
An Aquadag coating is applied to the inside
of the envelope to attract any secondary electrons emitted by the flourescent screen.
In the average electrostatic -deflection CR
tube the spot will be fairly well centered if all
four deflection plates are returned to the po-

tential of the second anode (ground). How ever, for accurate centering and to permit moving the entire trace either horizontally or
vertically to permit display of a particular
waveform, horizontal and vertical centering
controls usually are provided on the front of
the oscilloscope.
After the spot is once centered, it is necessary only to apply a positive or negative voltage (with respect to ground) to one of the
ungrounded or "free" deflector plates in order
to move the spot. If the voltage is positive
with respect to ground, the beam will be
attracted toward that deflector plate, while if
negative the beam and spot will be repulsed.
The amount of deflection is directly proportional to the voltage (with respect to ground)
that is applied to the free electrode.
With the larger- screen higher -voltage tubes
it becomes necessary to place deflecting voltage on both horizontal and both vertical plates.
This is done for two reasons: First, the amount
of deflection voltage required by the high-

www.americanradiohistory.com

Vacuum

86

Tube

FIRST

ANODE

\

THE

DEFLECTION COILS

. 11111
SECOND

(ADYADA.,Ï

_

/CONTROL

GRID

RADIO

T-TERMINAL

FOCUS COIL

BASE

Principles

®

10

_ _

1tI1.-o
R_iirr-

-v

ECEETFOÑR[AM
FLUORESCENT SCREEN

-

ID)

/CATHODE (R)

A

R

Figure 29
TYPICAL ELECTROMAGNETIC
CATHODE -RAY TUBE

voltage tubes is so great that a transmitting
tube operating from a high voltage supply
would be required to attain this voltage without distortion. By using push -pull deflection
with two tubes feeding the deflection plates,
the necessary plate supply voltage for the deflection amplifier is halved. Second, a certain
amount of de- focussing of the electron stream
is always present on the extreme excursions in
deflection voltage when this voltage is applied
only to one deflecting plate. When the deflecting voltage is fed in push -pull to both
deflecting plates in each plane, there is no defocussing because the average voltage acting
on the electron stream is zero, even though the
net voltage (which causes the deflection)
acting on the stream is twice that on either
plate.
The fact that the beam is deflected by a
magnetic field is important even in an oscilloscope which employs a tube using electrostatic deflection, because it means that precautions must be taken to protect the tube from
the transformer fields and sometimes even the
earth's magnetic field. This normally is done
by incorporating a magnetic shield around the
tube and by placing any transformers as far
from the tube as possible, oriented to the position which produces minimum effect upon the
electron stream.
The
electromagnetic
cathode -ray tube allows
greater definition than
does the electrostatic tube. Also, electromagnetic definition has a number of advantages when a rotating radial sweep is required
to give polar indications.
The production of the electron beam in an
electromagnetic tube is essentially the same
as in the electrostatic tube. The grid structure
is similar, and controls the electron beam in
an identical manner. The elements of a typical
electromagnetic tube are shown in figure 29.
The focus coil is wound on an iron core which
may be moved along the neck of the tube to
focus the electron beam. For final adjustment,
Construction of Electro-

magnetic CRT

.1m

Jew

.1n1R

Figure 30
Two pairs of coils arranged for electromagnetic deflection in two directions.

the current flowing in the coil may be varied.
A second pair of coils, the deflection coils
are mounted at right angles to each other
around the neck of the tube. In some cases,
these coils can rotate around the axis of the
tube.
Two anodes are used for accelerating the
electrons from the cathode to the screen. The
second anode is a graphite coating (Aquadag)
on the inside of the glass envelope. The function of this coating is to attract any secondary
electrons emitted by the flourescent screen,
and also to shield the electron beam.
In some types of electromagnetic tubes, a
first, or accelerating anode is also used in
addition to the Aquadag.
Electromagnetic
Deflection

magnetic field will deflect
an electron beam in a direction which is at right angles
to both the direction of the field and the direction of motion of the beam.
In the general case, two pairs of deflection
coils are used (figure 30). One pair is for
horizontal deflection, and the other pair is for
vertical deflection. The two coils in a pair
are connected in series and are wound in such
directions that the magnetic field flows from
one coil, through the electron beam to the
other coil. The force exerted on the beam by
the field moves it to any point on the screen
by application of the proper currents to these
A

coils.

The human eye retains an image
for about one - sixteenth second
after viewing. In a CRT, the spot can be
moved so quickly that a series of adjacent
spots can be made to appear as a line, if the
beam is swept over the path fast enough. As
The Trace

www.americanradiohistory.com

HANDBOOK

Gas

long as the electron beam strikes in a given
place at least sixteen times a second, the
spot will appear to the human eye as a source
of continuous light with very little flicker.

-

least five types of luminescent screen materials
are commonly available on
the various types of CR tubes commercially
available. These screen materials are called
phosphors; each of the five phosphors is best
suited to a particular type of application. The
P -1 phosphor, which has a green flourescence
with medium persistence, is almost invariably
used for oscilloscope tubes for visual observaScreen Materials

"Phosphors"

At

tion. The P -4 phosphor, with white fluorescence and medium persistence, is used on
television viewing tubes ( "Kinescopes "). The
P -5 and P -11 phosphors, with blue fluorescence and very short persistence, are used
primarily in oscilloscopes where photographic
recording of the trace is to be obtained. The
P -7 phosphor, which has a blue flash and a
long -persistence greenish -yellow persistence,
is used primarily for radar displays where
retention of the image for several seconds
after the initial signal display is required.
4 -9

Gas Tubes

The space charge of electrons in the vicinity
of the cathode in a diode causes the plate -tocathode voltage drop to be a function of the
current being carried between the cathode and
the plate. This voltage drop can be rather high
when large currents are being passed, causing
a considerable amount of energy loss which
shows up as plate dissipation.

Mercury Vapor
Tubes

Tubes

87

Mercury -vapor tubes, although
very widely used, have the

disadvantage that they must be
operated within a specific temperature range
(25° to 70°C.) in order that the mercury vapor
pressure within the tube shall be within the
proper range. If the temperature is too low,
the drop across the tube becomes too high
causing immediate overheating and possible
damage to the elements. If the temperature is
too high, the vapor pressure is too high, and
the voltage at which the tube will "flash back"
is lowered to the point where destruction of
the tube may take place. Since the ambient
temperature range specified above is within
the normal room temperature range, no trouble
will be encountered under normal operating
conditions. However, by the substitution of
xenon gas for mercury it is possible to produce a rectifier with characteristics comparable
to those of the mercury -vapor tube except that
the tube is capable of operating over the range
from approximately -70° to 90° C. The 3B25
rectifier is an example of this type of tube.
If a grid is inserted between the cathode and plate of a mercury -vapor
gaseous- conduction rectifier, a negative potential placed upon the added element
will increase the plate -to- cathode voltage drop
required before the tube will ionize or "fire."
The potential upon the control grid will have
no effect on the plate -to- cathode drop after the
tube has ionized. However, the grid voltage
may be adjusted to such a value that conduction will take place only over the desired
portion of the cycle of the a-c voltage being
impressed upon the plate of the rectifier.
Thyratron
Tubes

Voltage Regulator In a glow -discharge gas tube
Tubes
the voltage drop across the

electrodes remains constant

The negative space charge can
be neutralized by the presence
of the proper density of positive
ions in the space between the cathode and
anode. The positive ions may be obtained by
the introduction of the proper amount of gas or
a small amount of mercury into the envelope of
the tube. Then the voltage drop across the
tube reaches the ionization potential of the
gas or mercury vapor, the gas molecules will
become ionized to form positive ions. The
positive ions then tend to neutralize the space
charge in the vicinity of the cathode. The voltage drop across the tube then remains constant
at the ionization potential of the gas up to a
current drain equal to the maximum emission
capability of the cathode. The voltage drop
varies between 10 and 20 volts, depending
upon the particular gas employed, up to the
maximum current rating of the tube.
Action of
Positive Ions

over a wide range of current passing through
the tube. This property exists because the
degree of ionization of the gas in the tube
varies with the amount of current passing
through the tube. When a large current is
passed, the gas is highly ionized and the
internal impedance of the tube is low. When a
small current is passed, the gas is lightly
ionized and the internal impedance of the tube
is high. Over the operating range of the tube,
the product (IR) of the current through the tube
and the internal impedance of the tube is very
nearly constant. Examples of this type of tube
are VR -150, VR -105 and the old 874.
Vacuum tubes are grouped into
three major classifications:
commercial, ruggedized, and
premium (or reliable). Any one of these three
groups may also be further classified for
Vacuum Tube

Classification

www.americanradiohistory.com

88

Tube

Vacuum

THE

Principles
100-

military duty (JAN classification). To qualify
for JAN classification, sample lots of the
particular tube must have passed special
qualification tests at the factory. It should not
be construed that a JAN-type tube is better
than a commercial tube, since some commercial
tests and specifications are more rigid than
the corresponding JAN specifications. The
JAN -stamped tube has merely been accepted
under a certain set of conditions for military
service.

IP=2.5IAA.
eo
60

L

0
20
0

0

10

20
EP

Ruggedized or
Premium Tubes

Radio tubes are being used in
increasing numbers for industrial applications, such as
computing and control machinery, and in aviation and marine equipment. When a tube fails
in a home radio receiver, it is merely inconvenient, but a tube failure in industrial applications may bring about stoppage of some vital
process, resulting in financial loss, or even
danger to life.
To meet the demands of these industrial
applications, a series of tubes was evolved
incorporating many special features designed
to ensure a long and pre- determined operating
life, and uniform characteristics among similar
tubes. Such tubes are known as ruggedized or
premium tubes. Early attempts to select reTRIODE PLATE

,

`FLUORESCENT ANODE

TRIODE GRID

RAY CONTROL
ELECTRODE
CATHODES

Figure 31
SCHEMATIC REPRESENTATION
OF "MAGIC EYE" TUBE

liable specimens of tubes from ordinary stock
tubes proved that in the long run the selected
tubes were no better than tubes picked at
random. Long life and ruggedness had to be
built into the tubes by means of proper choice
and 100% inspection of all materials used in
the tube, by critical processing inspection and
assembling, and by conservative ratings of the
tube.

Pure tungsten wire is used for heaters in
preference to alloys of lower tensile strength.
Nickel tubing is employed around the heater
wires at the junction to the stem wires to
reduce breakage at this point. Element structures are given extra supports and bracing.
Finally, all tubes are given a 50 hour test run
under full operating conditions to eliminate
early failures. When operated within their
ratings, ruggedized or premium tubes should
provide a life well in excess of 10,000 hours.
Ruggedized tubes will withstand severe
impact shocks for short periods, and will

RADIO

30
40
VOLTS)

50

60

Figure 32

AMPLIFICATION FACTOR OF TYPICAL MODE
TUBE DROPS RAPIDLY AS PLATE VOLTAGE
IS

DECREASED BELOW 20 VOLTS

operate under conditions of vibration for many
hours. The tubes may be identified in many
cases by the fact that their nomenclature includes a "W" in the type number, as in 807W,
5U4W, etc. Some ruggedized tubes are included
in the "5000" series nomenclature. The 5654
is a ruggedized version of the 6AK5, the 5692
is a ruggedized version of the 6SN7, etc.
4 -10

lectron

Miscellaneous Tube Types

electron -ray tube or magic eye
contains two sets of elements, one
of which is a triode amplifier and
the other a cathode -ray indicator. The plate of
the triode section is internally connected to
the ray- control electrode (figure 31), so that
as the plate voltage varies in accordance with
the applied signal the voltage on the ray -control
electrode also varies. The ray -control electrode
is a metal cylinder so placed relative to the
cathode that it deflects some of the electrons
emitted from the cathode. The electrons which
strike the anode cause it to fluoresce, or give
off light, so that the deflection caused by the
ray -control electrode, which prevents electrons
from striking part of the anode, produces a
wedge- shaped electrical shadow on the fluorescent anode. The size of this shadow is determined by the voltage on the ray -electrode. When
this electrode is at the same potential as the
fluorescent anode, the shadow disappears; if
the ray -electrode is less positive than the
anode, a shadow appears the width of which
is proportional to the voltage on the ray -electrode. Magic eye tubes may be used as tuning
indicators, and as balance indicators in electrical bridge circuits. If the angle of shadow is
calibrated, the eye tube may be used as a voltmeter where rough measurements suffice.
E

The

Ray Tubes

www.americanradiohistory.com

Miscellaneous

HANDBOOK

89

Ecz (VOLTS)

Ec1=,z.ev
Figure 33

CHARACTERISTIC CURVES OF 12AK5
SFACE- CHARGE TRIODE

Controlled
Warm -up

Tubes

Series heater strings are employed
in ac -dc radio receivers and television sets to reduce the cost,

size, and weight of the equipment.
Voltage surges of great magnitude occur in

series operated filaments because of variations
in the rate of warm -up of the various tubes.
As the tubes warm up, the heater resistance
changes. This change is not the same between
tubes of various types, or even between tubes
same type made by different manufacturers. Some 6 -volt tubes show an initial
surge as high as 9 -volts during warm -up, while
slow -heating tubes such as the 25BQ6 are
underheated during the voltage surge on the
6 -volt tubes.
the

of

Standardization of heater characteristics in
a new group of cubes designed for series heater
strings has eliminated this trouble. The new
tubes have either 600 ma. or 400 ma. heaters,
with a controlled warm -up time of approximately
11 seconds. The 5U8, 6CG7, and 12BH7 -A are
examples of controlled warm -up tubes.
Introduction of the 12 -volt ignition
system in American automobiles
Potential
has brought about the design of a
Tubes
series of tubes capable of operation
with a plate potential of 12 -14
volts. Standard tubes perform poorly at low
plate potentials, as the amplification factor
of the tube drops rapidly as the plate voltage
is decreased (figure 32). Contact potential
effects, and change of characteristics with
variations of filament voltage combine to make
operation at low plate potentials even more
Low

Plate

erratic.
By employing

and by altering the electrode geometry a series
of low voltage tubes has been developed by
Tung -Sol that effectively perform with all electrodes energized by a 12 -volt system. With a
suitable power output transistor, this makes
possible an automobile radio without a vibrator
power supply. A special space- charge tube
(12K5) has been developed that delivers 40
milliwatts of audio power with a 12 volt plate

supply (figure 33).
The increased number of imported
radios and high- fidelity equipment
have brought many foreign vacuum
tubes into the United States. Many of these
tubes are comparable to, or interchangeable
with standard American tubes. A complete
listing of the electrical characteristics and
hase connection diagrams of all general purpose tubes made in all tube -producing
countries outside the "Iron Curtain" is contained in the Radio Tube Vade Mecum (World's
Radio Tubes) available at most larger radio
parts jobbers or by mail from the publishers
of this Handbook. The Equivalent Tubes Vade
Mecum (World's Equivalent Tubes) gives all
replacement tubes for a given type, both exact
and near -equivalents (with points of difference detailed). (Data on TV and special purpose tubes if needed is contained in a companion volume Television Tubes Vade Mecum).
Foreign
Tubes

special processing techniques

www.americanradiohistory.com

CHAPTER FIVE

Transistors and
Semi -Conductors

One of the earliest detection devices used
in radio was the galena crystal, a crude example of a semiconductor. More modern examples of semiconductors are the copper -

5 -1

It has been previously stated that the electrons in an element having a large atomic
number are grouped into rings, each ring having a definite number of electrons. Atoms in
which these rings are completely filled are
called inert gases, of which helium and argon
are examples. All other elements have one or
more incomplete rings of electrons. If the incomplete ring is loosely bound, the electrons
may be easily removed, the element is called
metallic, and is a conductor of electric current.
If the incomplete ring is tightly bound, with
only a few missing electrons, the element is
called non - metallic and is an insulator of electric current. Germanium and silicon fall between these two sharply defined groups, and
exhibit both metallic and non -metallic characteristics. Pure germanium or silicon may be
considered to be a good insulator. The addition
of certain impurities in carefully controlled
amounts to the pure germanium will alter the
conductivity of the material. In addition, the
choice of the impurity can change the direction
of conductivity through the crystal, some impurities increasing conductivity to positive voltages, and others increasing conductivity to negative voltages.

oxide rectifier, the selenium rectifier and the
germanium diode. All of these devices offer
the interesting property of greater resistance
to the flow of electrical current in one direction than in the opposite direction. Typical
conduction curves for these semiconductors
are shown in Figure 1. The copper oxide rectifier action results from the function of a thin
film of cuprous oxide formed upon a pure copper disc. This film offers low resistance for
positive voltages, and high resistance for
negative voltages. The same action is observed in selenium rectifiers, where a film of
selenium is deposited on an iron surface.

s

1

1N3

CD0IIT*L DIODI

O

1

TYPICAL STATIC CHARACTERISTICS
00

w
w

o
I. I

1.1

I.1

-00

-.0

-SO

-

-20

0

0

5 -2
2

Mechanism of
Conduction

As indicated by their name, semiconductors
are substances which have a conductivity
intermediate between the high values observed
for metals and the low values observed for insulating materials. The mechanism of conduction in semiconductors is different from that

a

VOLTS

Figure

Atomic Structure of
Germanium and Silicon

lA

TYPICAL CHARACTERISTIC CURVE
OF SEMI -CONDUCTOR DIODE

90
www.americanradiohistory.com

Transistors

observed in metallic conductors. There exist
in semiconductors both negatively charged
electrons and positively charged particles,
called holes, which behave as though they
had a positive electrical charge equal in magnitude to the negative electrical charge on
the electron. These holes and electrons drift
in an electrical field with a velocity which is
proportional to the field itself:

SCHEMATIC REPRESENTATION

VAN

where

VAN

E

_

-1=011

ANODES

Color

CATB000
BEOI

M11

¡

l.

Calorr Bao ft

-- Wrba

TUBE. GERMANIUM. SILICON
AND SELENIUM DIODES

Figure
COMMON DIODE
AND MARKINGS
IN ABOVE

1

91

-B

COLOR CODES
ARE SHOWN

CHART

=

µnE

= drift velocity of hole
= magnitude of electric field
= mobility of hole

In an electric field the holes will drift in a
direction opposite to that of the electron and
with about one-half the velocity, since the
hole mobility is about one -half the electron
mobility. A sample of a semiconductor, such as
germanium or silicon, which is both chemically
pure and mechanically perfect will contain in it
approximately equal numbers of holes and electrons and is called an intrinsic semiconductor.
The intrinsic resistivity of the semiconductor
depends strongly upon the temperature, being
about 50 ohm /cm. for germanium at room
temperature. The intrinsic resistivity of silicon
is about 65,000 ohm /cm. at the same temperature.
If, in the growing of the semiconductor crystal, a small amount of an impurity, such as
phosphorous, arsenic or antimony is included
in the crystal, each atom of the impurity contributes one free electron. This electron is
available for conduction. The crystal is said
to be doped and has become electron- conductPe- Nb JUNCTION

PLASTIC CASE

b-PC JUNCTION

P - TYPE

N- TYPE

GERMANIUM
CRYSTAL LAYER

GERMANIUM
CRYSTAL LAYER
COLLECTOR

EMITTER

LI

Nb
P

.MS'

BASE CONNECTION

SMALL 3 -PIN
BASE

L

Jw

y1!

o

o

Pt

4-

Z

.320
ASE CONNECTION

EMITTER

COLLECTOR

Figure 2A
CUT -AWAY VIEW OF JUNCTION
TRANSISTOR, SHOWING PHYSICAL
ARRANGEMENT

Figure 2B
PICTORIAL EQUIVALENT OF
P -N -P JUNCTION TRANSISTOR

www.americanradiohistory.com

SIGN Z

92

THE RADIO

Transistors and Semi- Conductors
EMITTER

COLLECTOR

EMITTER

COLLECTOR

BASE-I

BISE
TRANSISTOR OR
POINT CONTACT TRANSISTOR

N-P-N TRANSISTOR

P -N -P

Figure 4
ELECTRICAL SYMBOLS
FOR TRANSISTORS
Figure 3
CONSTRUCTION DETAIL OF A
POINT CONTACT TRANSISTOR

ing in nature and is called N (negative) type
germanium. The impurities which contribute
electrons are called donors. N -type germanium
has better conductivity than pure germanium in
one direction, and a continuous stream of electrons will flow through the crystal in this direction as long as an external potential of the
correct polarity is applied across the crystal.
Other impurities, such as aluminum, gallium or indium add one hole to the semiconducting crystal by accepting one electron for
each atom of impurity, thus creating additional
holes in the semiconducting crystal. The material is now said to be hole- conducting, or P
(positive) type germanium. The impurities
which create holes are called acceptors. P -type
germanium has better conductivity than pure
germanium in one direction. This direction is
opposite to that of the N -type material. Either
the N -type or the P -type germanium is called
extrinsic conducting type. The doped materials
have lower resistivities than the pure materials,
and doped semiconductors in the resistivity
range of .01 to 10 ohm /cm. are normally used
in the production of transistors.

5-3

The Transistor

In the past few years an entire new technology has been developed for the application
of certain semiconducting materials in production of devices having gain properties. These
gain properties were previously found only in
vacuum tubes. The elements germanium and
silicon are the principal materials which exhibit the proper semiconducting properties permitting their application in the new amplifying devices called transistors. However,
other semiconducting materials, including the
compounds indium antimonide and lead sulfide
have been used experimentally in the production of transistors.

Types of Transistors

There

are two basic
types of transistors, the
point-contact type and the junction type (figure 2) . Typical construction detail of a pointcontact transistor is shown in Figure 3, and
the electrical symbol is shown in Figure 4. The
emitter and collector electrodes make contact
with a small block of germanium, called the
base. The base may be either N -type or P -type
germanium, and is approximately .05" long
and .03" thick. The emitter and collector electrodes are fine wires, and are spaced about
.005" apart on the germanium base. The complete assembly is usually encapsulated in a
small, plastic case to provide ruggedness and
to avoid contaminating effects of the atmosphere. The polarity of emitter and collector
voltages depends upon the type of germanium
employed in the base, as illustrated in figure 4.
The junction transistor consists of a piece
of either N -type or P -type germanium between
two wafers of germanium of the opposite type.
Either N -P -N or P -N -P transistors may be
made. In one construction called the grown
crystal process, the original crystal, grown
from molten germanium or silicon, is created
in such a way as to have the two closely spaced
junctions imbedded in it. In the other construction called the fusion process, the crystals
are grown so as to make them a single conductivity type. The junctions are then produced by fusing small pellets of special metal
alloys into minute plates cut from the original
crystal. Typical construction detail of a junction
transistor is shown in figure 2A.
The electrical schematic for the P -N -P junction transistor is the same as for the pointcontact type, as is shown in figure 4.
Transistor Action

Presently available types of
transistors have three essential actions which collectively are called
transistor action. These are: minority carrier
injection, transport, and collection. Figure 2B
shows a simplified drawing of a P-N -P junction -type transistor, which can illustrate this

www.americanradiohistory.com

HANDBOOK

Transistors

collective action. The P -N -P transistor consists of a piece of N -type germanium on opposite sides of which a layer of P -type material has been grown by the fusion process.
Terminals are connected to the two P- sections
and to the N -type base. The transistor may be
considered as two P -N junction rectifiers
p!aced in close juxaposition with a semi conduction crystal coupling the two rectifiers
together. The left -hand terminal is biased in
the forward (or conducting) direction and is
called the emitter. The right -hand terminal is
biased in the back (or reverse) direction and
is called the collector The operating potentials
are chosen with respect to the base terminal,
which may or may not be grounded. If an
N -P -N transistor is used in place of the P -N -P,
the operating potentials are reversed.
The P.
Nb junction on the left is biased
in the forward direction and holes from the
P region are injected into the Nb region, producing therein a concentration of holes substantially greater than normally present in the
material. These holes travel across the base
region towards the collector, attracting neighboring electrons, finally increasing the available supply of conducting electrons in the
collector loop. As a result, the collector loop
possesses lower resistance whenever the emitter circuit is in operation. In junction transistors this charge transport is by means of
diffusion wherein the charges move from a
region of high concentration to a region of
lower concentration at the collector. The collector, biased in the opposite direction, acts
as a sink for these holes, and is said to collect them.
It is known that any rectifier biased in the
forward direction has a very low internal impedance, whereas one biased in the back direction has a very high internal impedance. Thus,
current flows into the transistor in a low impedance circuit, and appears at the output as
current flowing in a high impedance circuit.
The ratio of a change in collector current to
a change in emitter current is called the current
amplification, or alpha:

-

a

=

ie

= current amplification
= change in collector current
i. = change in emitter current

where a
ie

Values of alpha up to 3 or so may be obtained in commercially available point- contact
transistors, and values of alpha up to about
0.95 are obtainable in junction transistors.

93

Alpha Cutoff

The alpha cutoff frequency of
a transistor is that frequency
at which the grounded base
current gain has decreased to 0.7 of the gain
obtained at 1 kc. For audio transistors, the
alpha cutoff frequency is in the region of 0.7
Mc. to 1.5 Mc. For r -f and switching transistors, the alpha cutoff frequency may be 5 Mc.
or higher. The upper frequency limit of operation of the transistor is determined by the
small but finite time it takes the majority carrier to move from one electrode to another.
Frequency

Drift Transistors

As previously noted, the
signal current in a conventional transistor is transmitted across the
base region by a diffusion process. The transit
time of the carriers across this region is, therefore relatively long. RCA has developed a

technique for the manufacture of transistors
which does not depend upon diffusion for
transmission of the signal across the base region. Transistors featuring this new process are
known as drift transistors. Diffusion of charge
carriers across the base region is eliminated and
the carriers are propelled across the region by
a "built in electric field. The resulting reduction of transit time of the carrier permits drift
transistors to be used at much higher frequencies than transistors of conventional design.

The "built in" electric field is in the base
region of the drift transistor. This field is
achieved by utilizing an impurity density
which varies from one side of the base to the
other. The impurity density is high next to
the emitter and low next to the collector. Thus,
there are more mobile electrons in the region
near the emitter than in the region near the
collector, and they will try to diffuse evenly
throughout the base. However, any displacement of the negative charge leaves a positive
charge in the region from which the electrons
came, because every atom of the base material
was originally electrically neutral. The displacement of the charge creates an electric
field that tends to prevent further electron diffusion so that a condition of equilibrium is
reached. The direction of this field is such as
to prevent electron diffusion from the high
density area near the emitter to the low density
area near the collector. Therefore, holes entering the base will be accelerated from the emitter to the collector by the electric field. Thus
the diffusion of charge carriers across the base
region is augmented by the built -in electric
field. A potential energy diagram for a drift
transistor is shown in figure 5.

www.americanradiohistory.com

94

\
%
ri_
\
:i1,
r_=/

THE RADIO

Transistors and Semi -Conductors
DECREASING
POTENTIAL ENERGY
OF

S

MAJORITY CARRIER

h
WW

7

CC

6

34

REGION

u
I

EMITTER

CASE
REGION

I

I

ION

I

COLLECTOR

REGION

I

Ifi/Ciioill111E.1
IMTINEE:011101
o1 l'/.
'/_!=
S

DRIFT

u

M
IMI
.
,1\I/I
IIIPAlf
I/ICJ
1
10

I

I

DISTANCE

Figure

20

s0

40

50

COLLECTOR VOLTS

0

5

POTENTIAL ENERGY DIAGRAM
DRIFT TRANSISTOR (2N247)

FOR

5 -4

_Aiii/1111

Transistor
Characteristics

The transistor produces results that may be
comparable to a vacuum tube, but there is a
basic difference between the two devices. The
vacuum tube is a voltage controlled device
whereas the transistor is a current controlled
device. A vacuum tube normally operates with
its grid biased in the negative or high resistance direction, and its plate biased in the
positive or low resistance direction. The tube
conducts only by means of electrons, and has
its conducting counterpart in the form of the
N -P -N transistor, whose majority carriers are
also electrons. There is no vacuum tube equivalent of the P -N -P transistor, whose majority
carriers are holes.
The biasing conditions stated above provide
the high input impedance and low output impedance of the vacuum tube. The transistor is
biased in the positive or low resistance direction in the emitter circuit, and in the negative,
or high resistance direction in the collector
circuit resulting in a low input impedance
and a high output impedance, contrary to and
opposite from the vacuum tube. A comparison
of point-contact transistor characteristics and
vacuum tube characteristics is made in figure 6.
The resistance gain of a transistor is expressed as the ratio of output resistance to
input resistance. The input resistance of a
typical transistor is low, in the neighborhood
of 300 ohms, while the output resistance is
relatively high, usually over 20,000 ohms. For
a point- contact transistor, the resistance gain
is usually over 60.
The voltage gain of a transistor is the
product of alpha times the resistance gain,
and for a point- contact transistor is of the

FAlBM!lPA

25

so

75

100

125

ISO

175

200

PLATE VOLTS

Figure 6
COMPARISON OF POINT -CONTACT
TRANSISTOR AND VACUUM TUBE
CHARACTERISTICS

order of 3 X 60 = 180. A junction transistor
which has a value of alpha less than unity
nevertheless has a resistance gain of the order
of 2000 because of its extremely high output
resistance, and the resulting voltage gain is
about 1800 or so. For both types of transistors
the power gain is the product of alpha squared
times the resistance gain and is of the order
of 400 to 500.
The output characteristics of the junction
transistor are of great interest. A typical example is shown in figure 7. It is seen that the
junction transistor has the characteristics of
an ideal pentode vacuum tube. The collector
current is practically independent of the collector voltage. The range of linear operation
extends from a minimum voltage of about 0.2
volts up to the maximum rated collector voltage. A typical load line is shown, which illustrates the very high load impedance that
would be required for maximum power transfer. A grounded emitter circuit is usually used,
since the output impedance is not as high as
when a grounded base circuit is used.

www.americanradiohistory.com

HANDBOOK

Transistor Characteristics
d

95

le

COLLECTOR

EMITTER

CASE

VALUES

OF THE EQUIVALENT

CIRCUIT

POINT- CONTACT
-1 0 -0.S

0

+5

+10

+ 5

+20

ISTOR

+25

COLLECTOR VOLTS

Vs..iMA VC15V.)

The output characteristics of a typical point contact transistor are shown in figure 6. The
pentode characteristics are less evident, rind the
output impedance is much lower, with the
range of linear operation extending down to
a collector voltage of 2 or 3. Of greater practical interest, however, is the input characteristic curve with short -circuited, or nearly shortcircuited input, as shown in figure 8. It is
this point -contact transistor characteristic of
having a region of negative impedance that
lends the unit to use in switching circuits. The
transistor circuit may be made to have two,
one or zero stable operating points, depending
upon the bias voltages and the load impedance
used.
Equivalent Circuit
of a Transistor

As is known from net -

work theory, the small
signal performance of
any device in any network can be represented
by means of an equivalent circuit. The most

EMITTER MILLIAMPERES

(te)

Figure 8
EMITTER CHARACTERISTIC CURVE
FOR TYPICAL POINT CONTACT
TRANSISTOR

JUNCTION
ISTOR
IMA VCR SV.)

re -EMITTER

1O0ß

SOA

Cb -SASE

300A

SOOA

RESISTANCE

Figure 7
OUTPUT CHARACTERISTICS OF
TYPICAL JUNCTION TRANSISTOR

(LE

RESISTANCE
RESCSÁL10EOR

c4- CURRENT

AMPLIFICATION

20000A
2.0

1

MEGONM

0.57

Figure 9
LOW FREQUENCY EQUIVALENT
(Common Bose) CIRCUIT FOR POINT
CONTACT AND JUNCTION
TRANSISTOR

convenient equivalent circuit for the low frequency small signal performance of both pointcontact and junction transistors is shown in
figure 9. r., rN, and rT, are dynamic resistances
which can be associated with the emitter, base
and collector regions of the transistor. The
current generator aI., represents the transport
of charge from emitter to collector. Typical
values of the equivalent circuit are shown in
figure 9.
Transistor
Configurations

There are three basic transistor configurations: grounded
base connection, grounded
emitter connection, and grounded collector
connection. These correspond roughly to
grounded grid, grounded cathode, and grounded plate circuits in vacuum tube terminology
(figure 10) .
The grounded base circuit has a low input
impedance and high output impedance, and no
phase reversal of signal from input to output
circuit. The grounded emitter circuit has a
higher input impedance and a lower output
impedance than the grounded base circuit, and
a reversal of phase between the input and output signal occurs. This circuit usually provides
maximum voltage gain from a transistor. The
grounded collector circuit has relatively high
input impedance, low output impedance, and
no phase reversal of signal from input to output circuit. Power and voltage gain are both
low.

Figure 11 illustrates some practical vacuum
tube circuits, as applied to transistors.

www.americanradiohistory.com

Transistors and Semi -Conductors

96

GROUNDED EMITTER
CONNECTION

GROUNDED BASE
CONNECTION

THE RADIO

GROUNDED COLLECTOR

CONNECTION

Figure 10
COMPARISON OF BASIC VACUUM TUBE AND TRANSISTOR CONFIGURATIONS

5 -5

Transistor Circuitry

To establish the correct operating parameters
of the transistor, a bias voltage must be established between the emitter and the base. Since
transistors are temperature sensitive devices,
and since some variation in characteristics usually exists between transistors of a given type,
attention must be given to the bias system to

FLIP -FLOP COUNTER

R. F.

overcome these difficulties. The simple self -bias
system is shown in figure 12A. The base is
simply connected to the power supply through
a large resistance which supplies a fixed value
of base current to the transistor. This bias
system is extremely sensitive to the current
transfer ratio of the transistor, and must be adjusted for optimum results with each transistor.
When the supply voltage is fairly high and

OSCILLATOR

ONE -STAGE RECEIVER

RFC

CRYSTAL OSCILLATOR

BLOCKING OSCILLATOR

DIRECT -COUPLED AMPLIFIER

Figure 11

TYPICAL TRANSISTOR CIRCUITS

www.americanradiohistory.com

AUDIO AMPLIF ER

HANDBOOK

Transistor Circuitry

97

-E
E

BIAS

BIAS

LOAD
RESISTOR

RESISTOR

RESISTOR

LOAD
RESISTOR

LOAD

RESISTOR

R2= lo Re
Re = soo- +000
2
SO

+

e

O

n

Li
(REVERSE POLARITY
FOR NPN TRANSISTOR)

O

Figure 12
BIAS CONFIGURATIONS FOR TRANSISTORS.
The voltage divider system of C is recommended for general transistor use. Ratio of
establishes base bias, and emitter bias is provided by voltage drop across Re.
Battery Polarity is reversed for N -P -N transistors.

wide variations in ambient temperature do not
occur, the bias system of figure 12B may be
used, with the bias resistor connected from
base to collector. When the collector voltage
is high, the base current is increased, moving
the operating point of the transistor down the
load line. If the collector voltage is low. the
operating point moves upwards along the load
line, thus providing automatic control of the
base bias voltage. This circuit is sensitive to
changes in ambient temperature, and may permit transistor failure when the transistor is
operated near maximum dissipation ratings.
A better bias system is shown in figure 12C,
where the base bias is obtained from a voltage
divider, (R1, R2 ), and the emitter is forward
biased. To prevent signal degeneration, the
emitter bias resistor is bypassed with a large
capacitance. A high degree of circuit stability
is provided by this form of bias, providing the
emitter capacitance is of the order of 50 t+fd.
for audio frequency applications.
Audio Circuitry

A simple voltage amplifier
is shown in figure 13. Distabilization is employed in the

rect current
enitter circuit. Operating parameters for the

R1

/R:

amplifier are given in the drawing. In this case,
the input impedance of the amplifier is quite
low. When used with a high impedance driving source such as a crystal microphone a step
down input transformer should be employed
as shown in figure 13B. The grounded collector circuit of figure 13C provides a high input
impedance and a low output impedance, much
as in the manner of a vacuum tube cathode
follower.
The circuit of a two stage resistance coupled
amplifier is shown in figure 14A. The input
impedance is approximately 1100 ohms. Feedback may be placed around this amplifier from
the emitter of the second stage to the base of
the first stage, as shown in figure 14B. A
direct coupled version of the r -c amplifier is
shown in figure 14C. The input impedance is
of the order of 15,000 ohms, and an overall
voltage gain of 80 may be obtained with a
supply potential of 12 volts.
It is possible to employ N -P -N and P -N -P
transistors in complementary symmetry circuits
which have no equivalent in vacuum tube design. Figure 15A illustrates such a circuit. A
symmetrical push -pull circuit is shown in

-t2V

VOLTAGE GAIN = 80
INPUT IMPEDANCE L 1200

VOLTAGE GAI N

P -N

.

0 97

INPUT IMPEDANCE 'L 300 /l

11

Figure 13
VOLTAGE AMPLIFIERS

-P TRANSISTOR

11

A resistance coupled amplifier employing an inexpensive CK -722 transistor is shown in A. For use with
a high impedance crystal microphone, a step -down transformer matches the low input impedance of the
transistor, as shown in B. The grounded collector configuration of C provides an input impedance of

about 300,000 ohms.

www.americanradiohistory.com

98

THE RADIO

Transistors and Semi -Conductors

4.7N

6A/N'L

IOUF

R2

Figure 14

TWO STAGE TRANSISTOR AUDIO AMPLIFIERS

The feedback loop of B may be added to the r -c amplifier to reduce distortion, or to control the
audio response. A direct coupled amplifier is shown in C.

figure 15B. This circuit may be used to directly drive a high impedance loudspeaker,
eliminating the output transformer. A direct
coupled three stage amplifier having a gain
figure of 80 db is shown in figure 15C.
The transistor may also be used as a class
A power amplifier, as shown in figure 16A.
Commercial transistors are available that will
provide five or six watts of audio power when
operating from a 12 volt supply. The smaller
units provide power levels of a few milli watts. The correct operating point is chosen so
that the output signal can swing equally in the
positive and negative directions, as shown in
the collector curves of figure 16B.
The proper primary impedance of the output transformer depends upon the amount of
power to be delivered to the load:

RP

E:
2R.

The collector current bias is:

Ic-=

213"

E,

In a class A output stage, the maximum a -c

power output obtainable is limited to 0.5 the
allowable dissipation of the transistor. The
product I, E, determines the maximum collector
dissipation, and a plot of these values is shown
in figure 16B. The load line should always lie
under the dissipation curve, and should encompass the maximum possible area between the
axes of the graph for maximum output condition. In general, the load line is tangent to the
dissipation curve and passes through the supply
voltage point at zero collector current. The d -c
operating point is thus approximately one -half
the supply voltage.
The circuit of a typical push -pull class B
transistor amplifier is shown in figure 17A.
Push -pull operation is desirable for transistor
operation, since the even -order harmonics are
largely eliminated. This permits transistors to
be driven into high collector current regions
without distortion normally caused by non linearity of the collector. Cross -over distortion
is reduced to a minimum by providing a slight
forward base bias in addition to the normal
emitter bias. The base bias is usually less than
0.5 volt in most cases. Excessive base bias will
boost the quiescent collector current and thereby lower the overall efficiency of the stage.
2N78
NPN

PNP

NPN

2N78
NPN

2N77
PNP

+E

só_11

SPEAKER

O

O
Figure 15
COMPLEMENTARY SYMMETRY AMPLIFIERS.

N -P -N and P -N -P transistors may be combined in circuits which have no equivalent in vacuum tube
design. Direct coupling between cascaded stages using a single power supply source may be employed, as
in C. Impedance of power supply should be extremely low.

www.americanradiohistory.com

HANDBOOK

Transistor Circuitry

99

2N 187A
I MAX

MAXIMUM COLLECTOR
DISSIPATION (IC X EC)

Figure 16

TYPICAL CLASS -A
AUDIO POWER

OPERATING POINT

TRANSISTOR CIRCUIT.
The

correct operating point

is

:hosen so that output signal can
swing equally in a positive or
negative direction, without ex:ceding maximum collector dis-

The operating point of the class B amplifier is set on the I. =O axis at the point where
the collector voltage equals the supply voltage.
The collector to collector impedance of the

output transformer is:
Rc-.

=

2Er'
Po

In the class B circuit, the maximum a -c
power input is approximately equal to five
times the allowable collector dissipation of
each transistor. Power transistors, such as the
2N301 have collector dissipation ratings of
5.5 watts and operate with class B efficiency
of about 67%. To achieve this level of operation the heavy duty transistor relies upon efficient heat transfer from the transistor case
to the chassis, using the large thermal capacity
of the chassis as a heat sink. An infinite heat
sink may be approximated by mounting the
transistor in the center of a 6" x 6" copper or
aluminum sheet. This area may be part of a
'arger chassis.
The collector of most power transistors is
electrically connected to the case. For applications where the collector is not grounded a
thin sheet of mica may be used between the
case of the transistor and the chassis.
Power transistors such as the Philco T -1041
may be used in the common collector class B
a.7N

configuration (figure 17C) to obtain high
power output at very low distortions comparable with those found in quality vacuum tube
circuits having heavy overall feedback. In addition, the transistor may be directly bolted to
the chassis, assuming a negative grounded
power supply Power output is of the order of
10 watts, with about 0.5% total distortion.
Circuitry

Transistors may be used for
radio frequency work provided
the alpha cutoff frequency of the units is
sufficiently higher than the operating frequency. Shown in figure 18A is a typical i -f
amplifier employing an N -P -N transistor. The
collector current is determined by a voltage
divider on the base circuit and by a bias resistor in the emitter leg. Input and output are
coupled by means of tuned i -f transformers.
Bypass capacitors are placed across the bias
resistors to prevent signal frequency degeneration. The base is connected to a low impedance untuned winding of the input transformer, and the collector is connected to a tap
on the output transformer to provide proper
matching, and also to make the performance of
the stage relatively independent of variations
between transistors of the same type. With a
rate -grown N -P -N transistor such as the G.E.
2N293, it is unnecessary to use neutralization
to obtain circuit stability. When P -N -P alloy
R -F

2N225

V

T-104_1

12V.

ZP-soonc.T.

ZS'3000CT.

2Ec

Ec

COLLECTOR VOLTAGE

sipation.

z

ZS=
LOAD

BOO

n

LINE

200 MW
NO SIGNAL

OPERATING
POINT

J

2N109

-T

R1
COLLECTOR VOLTAGE

EC

SO

ADJUST Ri FOR
V. BASE BIAS

O.4

1

X

ICC
ICC

í13v.
- 0.3 AMP.
(NAB.)',.35A.

PO'

10 WATTS

Figure 17
CLASS -B AUDIO AMPLIFIER CIRCUITRY.
C
permits
the
The common collector circuit of
transistor to be bolted directly to the chassis for efficient
heat transfer from the transistor case to the chassis.

100

Transistors and Semi- Conductors
2N293
N

THE RADIO
2N135

PN

p

TO

e
I

OUT.

MIXER
OR

P.

OUT.

TO
MIXER

OR
CONVERTER

CONVERTER

T N

r9v

Figure 18
TRANSISTORIZED I -F AMPLIFIERS.

Typical

transistor must be neutralized because of high collector capacitance.
grown N -P -N transistor does not usually require external neutralizing circuit.

P -N -P

Rate

1N64

Figure 19

AUTOMATIC VOLUME
CONTROL CIRCUIT

ro

MIXER
OR
CONVERTER

FOR TRANSISTORIZED
I -F
AMPLIFIER.

transistors are used, it is necessary to neutralize the circuit to obtain stability (figure 18B).
The gain of a transistor i -f amplifier will
decrease as the emitter current is decreased.
This transistor property can be used to control
the gain of an i -f amplifier so that weak and
strong signals will produce the same audio
output. A typical i -f strip incorporating this
automatic volume control action is shown in
figure 19.
R -f transistors may be used as mixers or
autodyne converters much in the same manner
as vacuum tubes The autodyne circuit is shown
in figure 20. Transformer T, feeds back a

signal from the collector to the emitter causing oscillation. Capacitor C, tunes thé oscillator
circuit to a frequency 455 kc. higher than that
of the incoming signal. The local oscillator
signal is inductively coupled into the emitter
circuit of the transistor. The incoming signal
is resonated in T_ and coupled via a low impedance winding to the base circuit. Notice
that the base is biased by a voltage divider
circuit much the same as is used in audio frequency operation. The two signals are mixed
in this stage and the desired beat frequency of
455 kc. is selected by i -f transformer T1 and
passed to the next stage. Collector currents of
0.6 ma. to 0.8 ma. are common, and the local
oscillator injection voltage at the emitter is in
the range of 0.15 to 0.25 volts, r.m.s.
A complete receiver "front end" capable of
operation up to 23 Mc. is shown in figure 21.
The RCA 2N247 drift transistor is used for
the r -f amplifier (TRI ), mixer (TR2), and
high frequency oscillator (TR3) The 2N247
incorporates an interlead shield, cutting the
interlead capacitance to .003 q fd. If proper
shielding is employed between the tuned circuits of the r -f stage, no neutralization of the
stage is required. The complete assembly obtains power from a 9 -volt transistor battery.
Note that input and output circuits of the transistors are tapped at low impedance points on
the r -f coils to achieve proper impedance match.
.

Figure 20
THE AUTODYNE CONVERTER CIRCUIT
USING A 2N168A AS A MIXER.

www.americanradiohistory.com

HANDBOOK

RF

Transistor Circuitry

101

Figure 21
AMPLIFIER, MIXER,
AND OSCILLATOR
STAGES FOR

TRANSISTORIZED
HIGH FREQUENCY
RECEIVER. THE RCA
2N247 DRIFT
TRANSISTOR IS
CAPABLE OF
EFFICIENT OPERATION
UP TO 23 Mc.

i

L

Sufficient coupling of the proper
input and output
circuits of the transistor will permit oscillation up to and slightly above the
alpha cutoff frequency. Various forms of transistor oscillators are shown in figure 22. A
simple grounded emitter Hartley oscillator having positive feedback between the base and the
collector (22A) is compared to a grounded
base Hartley oscillator (22B) . In each case
the resonant tank circuit is common to the input and output circuits of the transistor. Self bias of the transistor is employed in both these
circuits A more sophisticated oscillator employing a 2N247 transistor and utilizing a
voltage divider -type bias system (figure 22C)
is capable of operation up to 50 Mc. or so.
The tuned circuit is placed in the collector,
with a small emitter -collector capacitor providing feedback to the emitter electrode.
A P -N -P and an N -P -N transistor may be
combined to form a complementary Hartley
oscillator of high stability ( figure 23). The
collector of the P -N -P transistor is directly
Transistor
Oscillators

phase between

coupled to the base of the N -P -N transistor,
and the emitter of the N -P -N transistor furnishes the correct phase reversal to sustain oscillation. Heavy feedback is maintained between the emitter of the P -N -P transistor and
the collector of N -P -N transistor. The degree
of feedback is controlled by R1. The emitter
resistor of the second transistor is placed at the
+9V.

PwP

2N247

1N81

P -N

1N81

ion

NPN

2N78

Figure 23
COMPLEMENTARY HARTLEY
OSCILLATOR

-P and N -P -N transistors

bility

form high sta-

oscillator. Feedback between P -N -P
emitter and N -P -N collector is controlled by
R,. 1N81 diodes are used as amplitude limiters. Frequency of oscillation is determined
by L, C, -C :.

RFC

RFC

Figure 22

TYPICAL TRANSISTOR OSCILLATOR CIRCUITS

A- Grounded

Emitter Hartley
Grounded Base Hartley
C -2N247 Oscillator Suitable for

B-

SO

Mc. operation.

www.americanradiohistory.com

102

THE RADIO

Transistors and Semi -Conductors
2N33

POiNT-CONTA.T
TRANS.STON

POINT -CONTACT TRANSISTOR

LE

CI.ARGN
PER OO

RFC

:E

V.

Figure 25
RELAXATION OSCILLATOR USING
POINT- CONTACT OR SURFACE

Figure 24
NEGATIVE RESISTANCE OF
POINT -CONTACT TRANSISTOR
PERMITS HIGH FREQUENCY
OSCILLATION (50 Mc) WITHOUT
WITHOUT NECESSITY OF
EXTERNAL FEEDBACK PATH.

BARRIER TRANSISTORS.

Relaxation
Oscillators

Transistors have almost unlimited use in relaxation acid R -C oscillator service. The negative re-

sistance characteristic of the point contact transistor make it well suited to such application.
Surface barrier transistors are also widely used
in this service, as they have the highest alpha
cutoff frequency among the group of -alphaless-than-unity- transistors. Relaxation oscillators used for high speed counting require transistors capable of operation at repetition rates
of 5 Mc. to 10 Mc.

center of the oscillator coil to eliminate loading of the tuned circuit.
Two germanium diodes are employed as
amplitude limiters, further stabilizing amplifier operation. Because of the low circuit impedances, it is permissible to use extremely
high -C in the oscillator tank circuit, effectively
limiting oscillator temperature stability to variations in the tank inductance.
The point- contact transistor exhibits negative input and output resistances over part of
its operaing range, due to its unique ability
to multiply the input current. This characteristic affords the use of oscillator circuitry having no external feedback paths ( figure 24).
A high impedance resonant circuit in the base
lead produces circuit instability and oscillation
at the resonant frequency of the L -C circuit.
Positive emitter bias is used to insure thermal
circuit stability.

A simple emitter controlled relaxation oscillator is shown in figure 25, together with
its operating characteristic. The emitter of the
transistor is biased to cutoff at the start of the
cycle (point I) The charge on the emitter capacitor slowly leaks to ground through the
emitter resistor, R1. Discharge time is determined by the time constant of RICI. When the
emitter voltage drops sufficiently low to permit
the transistor to reach the negative resistance
region (point 2) the emitter and collector resistances drop to a low value, and the collector
.

*E
e.

PO5ITiwE

TRi.,,ER

PJLSE

EPUL 5E

M

OUT

NPN

-

1/111

NP

P N P

lON

Figure 26
TRANSISTORIZED BLOCKING OSCILLATOR (A) AND ECCLES -JORDAN
BI- STABLE MULTIVIBRATOR (B).

High -alpha transistors must be employed in counting circuits to reduce effects of
storage time caused by transit lag in transistor base.

www.americanradiohistory.com

HANDBOOK

Transistor Circuitry

103

2n
PHONE`,

C1- 123LLF, .W.

M /LLER

# 2110

C2-

791/11F, PART OF C I
L1 - "LOOPSTIC K. COIL, J.W MILLER

La- OSCILLATOR

COIL, J

W.

I

Figure 28
SCHEMATIC, TRANSISTORIZED BROADCAST BAND (S00
DYNE RECEIVER.

"

L OOP ST ICI

COIL

Figure 27

"WRIST RADIO" CAN BE MADE
WITH LOOPSTICK, DIODE, AND
INEXPENSIVE CK -722 TRANSISTOR.
A TWENTY FOOT ANTENNA WIRE
WILL PROVIDE GOOD RECEPTION
IN STRONG SIGNAL AREAS.

current is limited only by the collector resistor,
R. The collector current is abruptly reduced
by the charging action of the emitter capacitor
CI (point 3), bringing the circuit back to the
original operating point. The "spike" of collector current is produced during the charging
period of C. The duration of the pulse and the
pulse repetition frequency (p.r.f.) are controlled by the values of C, R1, R_, and R.
Transistors may also be used as blocking
oscillators (figure 26A) . The oscillator may
be synchronized by coupling the locking signal
to the base circuit of the transistor. An oscillator of this type may be used to drive a flip flop circuit as a counter. An Eccles -Jordan
bi- stable flip -flop circuit employing surface barrier transistors may be driven between "off"
and "on" positions by an exciting pulse as
shown in figure 26B. The first pulse drives
the "on" transistor into saturation. This transistor remains in a highly conductive state until
the second exciting pulse arrives. The transistor does not immediately return to the cut -off

-

#2003

MILLER 4 2002

TI -4SS RC. I.F. TRANSFORMER,
T2 -455 PI C. F. TRANSFORMER,

J.W

MILLER2031

J.W. MILLER

2032

1600 KC.) SUPERHETERO-

state, since a time lapse occurs before the output waveform starts to decrease. This storage
time is caused by the transit lag of the minority
carriers in the base of the transistor. Proper circuit design and the use of high -alpha transistors can reduce the effects of storage time to a
minimum. Driving pulses may be coupled to
the multivibrator through steering diodes as
shown in the illustration.

5 -6

Transistor Circuits

With the introduction of the dollar transistor, many interesting and unusual experiments and circuits may be built up by the beginner in the transistor field. One of the most
interesting is the "wrist watch" receiver, illustrated in figure 27. A diode and a transistor
amplifier form a miniature broadcast receiver,
which may be built in a small box and carried
on the person. A single 1.5 -volt penlite cell
provides power for the transistor, and a short
length of antenna wire will suffice in the vicinity of a local broadcasting station.
A transistorized superhetrodyne for broadcast reception is shown in figure 28. No antenna is required, as a ferrite "loop-stick" is
used for the r -f input circuit of the 2N136
mixer transistor. A miniature magnetic "hearing aid" type earphone may be employed with
this receiver.
A simple phonograph

amplifier designed

for use with a high impedance crystal pickup
is shown in figure 29. Two stages of amplification using 2N109 transistors are used to
drive two 2N109 transistors in a class B con-

figuration. Approximately 200 milliwatts of

www.americanradiohistory.com

104

Transistors and Semi -Conductors
2N109
2N1O9
220N
CRYSTAL
PICKUP

f0

0

i5M

2UF

2N109
0

27M

K

Figure 29
HIGH GAIN, LOW DISTORTION AUDIO AMPLIFIER, SUITABLE FOR USE
WITH A CRYSTAL PICKUP. POWER OUTPUT IS 250 MILLIWATTS.

power may be obtained with a battery supply
of 12 volts. Peak current drain under maximum signal conditions is 40 ma.
Shown in figure 30 is an inexpensive and
compact 25 watt transistorized modulator suitable for mobile use with an automobile having
a 12 volt ignition system. This unit may be
used to modulate a 6146 r.f. amplifier stage
running at 400 volts and 125 milliamperes
plate power input. The two DS -501 power
transistors (Delco) are mounted on a heat
sink made of a 6" x 6" x 1/8" aluminum plate.
The components are mounted on the sink
which serves as the chassis. Output transformer

T1

-I50

T., consists of a 6.3 volt filament transformer

with the ends of the low voltage winding connected to the collectors of the output transistors. Resting modulator current is about 0.7
amperes, rising to nearly 2 amperes on full
modulation peaks.
The modulator should be positioned so that
motor heat and warm air is deflected from the
unit, or the efficiency of the aluminum heat
sink will be impaired and damage to the output transistors may result. A good location for
the unit is under the dash against the firewall.
Microphone gain may be adjusted by changing the value of the 100 ohm, 2 watt series
resistor.

Figure 30.
TRANSISTORIZED 25 WATT MOBILE MODULATOR.
ohm primary, 490 ohm secondary (center tap on primary not used), Thordarson

TR -S.

T2-400 ohm primary, 4 and 16 ohm secondary. Stancor TA -41.
T3-6.3 volt center tap, 3a. (See text.)
Note-Output transistors are insulated from heat-sink by Delco
(s: 1221264).

insulators

mica

R.F.

LOAD

NOTE:

C /ACV /T RETURNS ro
BATTERY TERMINAL.

CONTROL

-Nor-

CONTROL

CIRCUIT

+12.e

V.

www.americanradiohistory.com

-1z.e

V.

Zener Diodes
5 -7

Zener Diodes

The Zener Diode is a semiconductor device
that can be used as a constant voltage reference, or as a control element. Zener diodes
are available in ratings to 50 watts, with zener
voltages of approximately 4 volts to 200 volts.
The zener diode has electrical characteristics that are derived from a rectifying junction which operates at a reverse bias condition
not normally used. The zener knee (figure 31)
and constant voltage plateau are obtained
when this rectifying junction is back -biased
above the junction breakdown voltage. The
break from non -conductance to conductance is
very sharp. At applied voltages greater than
the breakdown point, the voltage drop across
the diode junction becomes essentially constant for a relatively wide range of currents.
This is the zener control region.
Thermal dissipation is obtained by mounting the zener diode to a heat sink composed
of a large area of metal having free access
to ambient air.

-

ZENER KNEE
MA.

CONSTANT
VOLTAGE

Mi

M

_

CURRENT
1.S

1S

2

VP (VOLTS)

u

REVERSe
CHARACTERISTIC

S=
{M

AM.

N

FIGURE

31

BETWEEN .ZENER KNEE- AND POINT OF MAXIMUM ZENER CUR RENT, THE ZENER VOLTAGE IS ESSENTIALLY CONSTANT AT SOVOLTS.

DIODE
2 VOLT

REG.
VOLT.

FIGURE 32
-ZENER DIODE FUNCTIONS AS
VOLTAGE REGULATOR OVER
RANGE OF CONSTANT VOLTAGE

PLATEAU.
B -TWO ZENER DIODES OF DIFFER

OS

1.0

MAX.ZENER

10 v

1LOUT

MM

iM=
MlAT 5

The zener diode may be employed as a shunt regulator
(figure 32A) in the manner
of a typical "VR- tube." Two zener diodes may
be employed in the circuit of figure 32B to
supply very low values of regulated voltage.
Two opposed zener diodes can be used to provide a.c. clipping of both halves of the cycle
(figure 32C) . Zener diodes may also be used
to protect meter movements as they provide a
very low resistance shunt across the movement when the applied voltage exceeds a
certain critical level.

IN ti

AMMO

REVERSE VOLTAGE
10
30
20

Applications

A

2 10

CHARACTERISTIC

Zener Diode

UNREG.
VOLT

qM/iI
MMMMM

,.

MUM=

105

-

ENT VOLTAGE CAN PROVIDE

SMALL REGULATED VOLTAGE.
C- OPPOSED ZENER DIODES CLIP
BOTH HALVES OF CYCLE OF A.C.
WAVE.

www.americanradiohistory.com

CHAPTER SIX

Vacuum Tube Amplifiers

6 -1

Vacuum Tube Parameters

Electrode Potentials

Symbols for
Vacuum -Tube
Parameters

E«

ep
eg

Ep
Eg

grid- screen
conversion

`Bp
Cpi,
Cm
Cou,

I,

fpm

ipmax
igmaz
Ip
Ig

factor

tube)

grid

grid voltage
cutoff

average

grid current
fundamental

maximum
maximum

grid current

Other Symbols

grid- cathode
grid

Pi

plate- cathode

input
output

grid

minimum
maximum

-- average plate current
-peak
plate
-instantaneous plate current
instantaneous
-- static
plate current
static
- plate
P.-plate
-- plate dissipation plus bias losses)
lb

Intere!ectrode Capacitances

C¡gk

-- instantaneous
plate potential
instantaneous
potential
instantaneous plate voltage
-positive instantaneous
voltage
static plate voltage
---static
bias
Electrode Currents

factor

mu

-

eco

Gm

Gc

grid supply voltage (a negative

epmin
egmp

Tube Constants

--transconductance
plate resistance
/tu transconductance(mixer
-- -plate capacitance
capacitance
capacitance
--- capacitance
(tetrode
capacitance (tetrode

-d-c

peak grid excitation voltage (1/2 total
peak -to -peak grid swing)
Epm -peak plate voltage (! i total peak -to -peak
plate swing)
Egm

involving vacuum -tube parameters, the following symbols
will be used throughout this book:

R,

plate supply voltage (a positive

quantity)
quantity)

As an assistance in simplify ing and shortening expressions

ft- amplification

-d-c

Ebb

The ability of the control grid of a vacuum
tube to control large amounts of plate power
with a small amount of grid energy allows the
vacuum tube to be used as an amplifier. It is
this ability of vacuum tube s to amplify an
extremely small amount of energy up to almost
any level without change in anything except
amplitude which makes the vacuum tube such
an extremely valuable adjunct to modern electronics and communication.

or pentode)
or pentode)

Pp
Pd

power input
power output

grid driving power (grid

106

www.americanradiohistory.com

current

grid current

Classes of Amplifiers

ST
-F-_,
r--

r--1
17-T-

Cw ---

Ccv:7:

-

I

.7.-.7.

CcKi_
:=.
L

Clan

CiN

:t:

-,Cour
I

__ _J

1

PENTODE OR TETRODE

TRIODE

Figure
STATIC
INTERELECTRODE
TANCES WITHIN A TRIODE,
OR TETRODE
1

CAPACIPENTODE,

107

is the grid -to -plate capacitance, and A is
the stage gain. This expression assumes that
the vacuum tube is operating into a resistive
load such as would be the case with an audio
stage working into a resistance plate load in
the middle audio range.
The more complete expression for the input
admittance (vector sum of capacitance and
resistance) of an amplifier operating into any
type of plate load is as follows:
Cgp

Input capacitance = Cgk

+ (1 + A

cos

(9) Cgp

Input resistance

135-grid dissipation
Np
9p
9g

R1

Z1

--load
-

plate efficiency(expressed as a decimal)
one -half angle of plate current flow
one -half angle of grid current flow

resistance

load impedance

A

The relationships between certain of the electrode potentials
and currents within a vacuum
tube are reasonably constant under specified

alone
angle of the plate load impedance, positive for inductive
loads, negative for capacitive

Vacuum-Tube

conditions of operation. These relationships
are called vacuum -tube constants and are
listed in the data published by the manufacturers of vacuum tubes. The defining equations
for the basic vacuum -tube constants are given
in Chapter Four.
Interelectrode
The values of interelectrode
Capacitances and capacitance published in
Miller Effect
vacuum -tube tables are the
static values measured, in
the case of triodes for example, as shown in
figure 1. The static capacitances are simply
as shown in the drawing, but when a tube is
operating as amplifier there is another consideration known as Miller Effect which causes
the dynamic input capacitance to be different
from the static value. The output capacitance
of an amplifier is essentially the same as the
static value given in the published tube tables.
The grid -to-plate capacitance is also the same
as the published static value, but since the
Cgp acts as a small capacitance coupling energy back from the pl ate circuit to the grid
circuit, the dynamic input capacitance is equal
ro the static value plus an amount (frequently
much greater in the case of a triode) determined by the gain of the stage, the plate load
impedance, and the Cgp feedback capacitance.
The total value for an audio amplifier stage
can be expressed in the following equation:

(dynamic) _

(

static)

+

(A

+ 1) Cgp

where Co is the grid -to- cathode capacitance,

O

= phase

O

Constants

sin

Where: Cgk = grid -to- cathode capacitance
Cgp = grid -to-plate capacitance
A
= voltage amplification of the tube

It can be seen from the above that if the
plate load impedance of the stage is capacitive or inductive, there will be a resistive component in the input admittance of the stage.
The resistive component of the input admittance will be positive (tending to load the
circuit feeding the grid) if the load impedance
of the plate is capacitive, or it will be negative
(tending to make the stage oscillate) if the
load impedance of the plate is inductive.
Neutralization
of Interelectrode
Capacitance

Neutralization of the effects
of interelectrode capacitance
is employed most frequently

in the case of radio frequency power amplifiers. Before the introduction of the tetrode and pentode tube, triodes
were employed as neutralized Class A amplifiers in receivers. This practice has been
largely superseded in the present state of the
art through the use of tetrode and pentode
tubes in which the Cge or feedback capacitance has been reduced to such a low value
that neutralization of its effects is not necessary to prevent oscillation and instability.

6 -2

Classes and Types of
Vacuum -Tube Amplifiers

Vacuum -tube amplifiers are grouped into
various classes and sub -classes according to
the type of work they are intended to perform.
The difference between the various classes is
determined primarily by the value of average
grid bias employed and the maximum value of

www.americanradiohistory.com

108
the

grid.

Vacuum

Tube

Amplifiers

THE

exciting signal to he impressed upon the
A Class A amplifier is an amplifier
biased and supplied with excitation

Class A

Amplifier

of such amplitude that plate current flows continuously (360° of the exciting
voltage waveshape) and grid current does not
flow at any time. Such an amplifier is normally
operated in the center of the grid- voltage
plate- current transfer characteristic and gives
an output waveshape which is a substantial
replica of the input waveshape.
Class Al
Amplifier

This is another term applied to the
Class A amplifier in which grid
current does not flow over any
portion of the input wave cycle.
This is

a Class A amplifier operated under such conditions that the
grid is driven positive over a portion of the input voltage cycle, but plate current still flows over the entire cycle.

Class A2
Amplifier

Class AB1 This is an amplifier operated under
Amplifier
such conditions of grid bias and
exciting voltage that plate current
flows for more than one-half the input voltage
cycle but for less than the complete cycle. In
other words the operating angle of plate current flow is appreciably greater than 180° but

less than

360°. The suffix 1 indicates that grid
current does not flow over any portion of the
input cycle.

Class AB2 amplifier is operated
under essentially the same conditions of grid bias as the Class
AB t amplifier mentioned above, but the exciting voltage is of such amplitude that grid current flows over an appreciable portion of the
input wave cycle.

Class AB2

A

Amplifier

amplifier is biased sub stantially to cutoff of plate current
(without exciting voltage) so that
plate current flows essentially over one -half
Class

B

A Class B

Amplifier

the input voltage cycle. The operating angle

of plate current flow is essentially 180 °. The
Class B amplifier is almost always excited
to such an extent that grid current flows.

Class C amplifier is biased to a
Amplifier value greater than the value required for plate current cutoff and
is excited with a signal of such amplitude
that grid current flows over an appreciable
period of the input voltage waveshape. The
angle of plate current flow in a Class C amplifier is appreciably less than 180 °, or in
other words, plate current flows appreciably
Class

RADIO

C

A

Figure 2
TYPES OF BIAS SYSTEMS
A

B

-

C -

Grid bias
Cathode bias
Grid leak bias

less than one -half the time. Actually, the conventional operating conditions for a Class C
amplifier are such that plate current flows for
120° to 150° of the exciting voltage waveshape.

There are three general types of
amplifier circuits in use. These
types are classified on the basis
of the return for the input and output circuits.
Conventional amplifiers are called cathode return amplifiers since the cathode is effectively
grounded and acts as the common return for
both the input and output circuits. The second
type is known as a plate return amplifier or
cathode follower since the plate circuit is effectively at ground for the input and output
signal voltages and the output voltage or
power is taken between cathode and plate. The
third type is called a grid -return or grounded grid amplifier since the grid is effectively at
ground potential for input and output signals
and output is taken between grid and plate.
Types of

Amplifiers

6 -3

Biasing Methods

The difference of potential between grid and
cathode is called the grid bias of a vacuum
tube. There are three general methods of
providing this bias voltage. In each of these
methods the purpose is to establish the grid
at a potential with respect to the cathode
which will place the tube in the desired operating condition as determined by its charac-

teristics.

Grid bias may be obtained from a source of
voltage especially provided for this purpose,
as a battery or other d -c power supply. This
method is illustrated in figure 2A, and is
known as fixed bias.
A second biasing method is illustrated in
figure 2B which utilizes a cathode resistor
across which an IR drop is developed as a
result of plate current flowing through it. The

www.americanradiohistory.com

Amplifier

HANDBOOK
cathode of the tube is held at a positive potential with respect to ground by the amount of
the IR drop because the grid is at ground potential. Since the biasing voltage depends
upon the flow of plate current the tube cannot
be held in a cutoff condition by means of the
cat bode bias voltage developed across the
cathode resistor. The value of this resistor is
determined by the bias required and the plate
current which flows at this value of bias, as
found from the tube characteristic curves.
A capacitor is shunted across the bias resistor
to provide a low impedance path to ground for
the a -c component of the plate current which
results from an a-c input signal on the grid.
The third method of providing a biasing
voltage is shown in figure 2C, and is called
grid -leak bias. During the portion of the input
cycle which causes the grid to be positive
with respect to the cathode, grid current flows
from cathode to grid, charging capacitor C,.
When the grid draws current, the grid -to- cathode
resistance of the tube drops from an infinite
value to a very low value, on the order of
1,000 ohms or so, making the charging time
constant of the capacitor very short. This enables Cs to charge up to essentially the full
value of the positive input voltage and results
in the grid (which is connected to the low potential plate of the capacitor) being held essentially at ground potential. During the negative swing of the input signal no grid current
flows and the discharge path of Cg is through
the grid resistance which has a value of
500,000 ohms or so. The discharge time constant for C5 is, therefore, very long in comparison to the period of the input signal and
only a small part of the charge on C5 is lost.
Thus, the bias voltage developed by the discharge of Cs is substantially constant and the
grid is not permitted to follow the positive
portions of the input signal.

Distortion in Amplifiers

6 -4

There are three main types of distortion that
may occur in amplifiers: frequency distortion,
phase distortion and amplitude distortion.

distortion may occur
when some frequency components
of a signal are amplified more than
Frequency distortion occurs at low

Frequency

Distortion

Distortiob

109

OUTPUT
SIGNAL

Figure

3

Illustration of the effect of phase distortion
on input wave containing o third harmonic
signal

two stage amplifier. Although the amplitudes
of both components are amplified by identical
ratios, the output waveshape is considerably
different from the input signal because the
phase of the third harmonic signal has been
shifted with respect to the fundamental signal.
This phase shift is known as phase distortion,
and is caused principally by the coupling circuits between the stages of the amplifier.
Most coupling circuits shift the phase of a
sine wave, but this has no effect on the shape
of the output wave. However, when a complex
wave is passed through the same coupling
circuit, each component frequency of the waveshape may be shifted in phase by a different
amount so that the output wave is not a faithful reproduction of the input waveshape.
a

Amplitude
Distortion

If

a signal is passed through a vac uum tube that is operating on any

non -linear part of its characteristic,
amplitude distortion will occur. In such a region, a change in grid voltage does not result
in a change in plate current which is directly
proportional to the change in grid voltage. For
example, if an amplifies is excited with a signal that overdrives the tubes, the resultant
signal is distorted in amplitude, since the
tubes operate over a non -linear portion of their

characteristic.

Frequency

others.
frequencies if coupling capacitors between
stages are too small, or may occur at high frequencies as a result of the shunting effects of
the distributed capacities in the circuit.
Phase

Distortion

input signal con sisting of a fundamental and a
third harmonic is passed through
In

figure

3

an

6 -5

Resistance Capacitance Coupled
Audio -Frequency Amplifiers

Present practice in the design of audio-frequency voltage amplifiers is almost exclusively
to use resistance -capacitance coupling between the low -level stages. Both triodes and

www.americanradiohistory.com

1

1

Vacuum

0

Tube

A mp

l

i

fie

T H E

r s

Figure 4
CIRCUIT FOR RESISTANCE CAPACITANCE COUPLED TRIODE AMSTANDARD

PLIFIER STAGE

R A D

The voltage gain per stage of
a resistance -capacitance coupled triode amplifier can be calculated with the aid of the equivalent circuits
and expressions for the mid -frequency, high frequency, and low- frequency range given in
figure 5.
A triode R -C coupled amplifier stage is
normally operated with values of cathode resistor and plate load resistor such that the
actual voltage on the tube is approximately
one -half the d -c plate supply voltage. To
per Stage

will

be

are used; triode amplifier stages
discussed first.

R -C

Coupled
Triode Stages

4 illustrates the stand circuit for a resistance -

Figure
and

capacitance coupled amplifier

stage utilizing a triode tube with cathode bias.
In conventional audio-frequency amplifier design such stages are used at medium voltage

G

E

A_

A)

RP

RL RG

(RL+RC)+RL

RG

11EG

MID FREQUENCY RANGE

CGN

(DYNAMIC,
NEXT STAGE)

L=-LEG

-

A HIGH FREE).
A MID FREE).

1

Ni

1+ (REQ /XS)2
RL

R CO

1+

HIGH FREQUENCY RANGE
Xs

G

E=

-L

O

levels (from 0.01 to 5 volts peak on the grid
of the tube) and use medium -p triodes such
as the 6J5 or high -p triodes such as the 6SF5
or 6SL7 -GT. Normal voltage gain for a single
stage of this type is from 10 to 70, depending
upon the tube chosen and its operating conditions. Triode tubes are normally used in the
last voltage amplifier stage of an R -C amplifier since their harmonic distortion with large
output voltage (25 to 75 volts) is less than
with a pentode tube.
Voltage Gain

pentodes

I

A LOW FREQ.
A MID FREQ.

'

RL

RL
Rn

RG

2TTF (CPN+CGN (orNAMlc)

=

1+ (XC

/R)2

EG

Xc

-

R

= RG+

1

2 TTFCC

LOW FREQUENCY RANGE

RL RP
RL+ RP

Figure 5
Equivalent circuits and gain equations for a triode R -C coupled amplifier stage. In using these
equations, be sure to select the values of mu and RP which are proper for the static current and
voltages with which the tube will operate. These values may be obtained from curves published
in the RCA Tube Handbook RC -16.

www.americanradiohistory.com

'7IANDBOOK

R

-C

Amplifiers

Coupled

111

such as the 6SJ7. Normal voltage gain for a
stage of this type is from 60 to 250, depending upon the tube chosen and its operating
conditions. Pentode tubes are ordinarily used
the first stage of an R -C amplifier where the

high gain which they afford is of greatest advantage and where only a small voltage output
is required from the stage.
Figure

6

CIRCUIT FOR RESISTANCE
CAPACITANCE COUPLED PENTODE AMSTANDARD

PLIFIER STAGE

assist the designer of such stages, data on
operating conditions for commonly used tubes
is published in the RCA Tube Handbook RC -16.
It is assumed, in the case of the gain equations
of figure 5, that the cathode by -pass capacitor,
Ck, has a reactance that is low with respect
to the cathode resistor at the lowest frequency
to be passed by the amplifier stage.
Coupled
Pentode Stages
R -C

6 illustrates the stand circuit for a resistance -

Figure
and

capacitance coupled pentode
amplifier stage. Cathode bias is used and the
screen voltage is supplied through a dropping
resistor from the plate voltage supply. In conventional audio -frequency amplifier design
such stages are normally used at low voltage
levels (from 0.00001 to 0.1 volts peak on the
grid of the tube) and use moderate -Gm pentodes

The voltage gain per stage of a resistance capacitance coupled pentode amplifier can be
calculated with the aid of the equivalent circuits and expressions for the mid -frequency,
high- frequency, and low- frequency range given
in figure 7.
To assist the designer of such stages, data
on operating conditions for commonly used
types of tubes is published in the RCA Tube
Handbook RC -16. It is assumed, in the case of
the gain equations of figure 7, that the cathode
by -pass capacitor, Ck, has a reactance that is
low with respect to the cathode resistor at the
lowest frequency to be passed by the stage. It
is additionally assumed that the reactance of
the screen by -pass capacitor Cd, is low with
respect to the screen dropping resistor, Rd, at
the lowest frequency to be passed by the amplifier stage.
Cascade Voltage
Amplifier Stages

When voltage amplifier stages
are operated in such a manner

that the output voltage of the

first is fed to the grid of the second, and so
forth, such stages are said to be cascaded.
The total voltage gain of cascaded amplifier

t= -GMEc

c

A

=

GM REO

RL

REO

Figure 7
Equivalent circuits and
gain equations for a pentode R -C coupled amplifier
stage. In using these equations be sure to select the
values of Gm and Rp which
are proper for the static
currents and voltages with
which the tube will operate. These values may be
obtained from curves published in the RCA Tube

Re

RG
MID FREQUENCY RANGE

I= -GMEc
A HIGH FRED.

AMID FOtO.
R CO -

1+

HIGH FREQUENCY RANGE

Xs'

}(REO /Xs)2

RL

1

Rc+RP

277F (CPK+CGK (DYNAMIC)

A LOW FREQ. _

Handbook RC -16.

A MID

LOW FREQUENCY RANGE

Xc
R =

www.americanradiohistory.com

I+(XCF R)2

FOCO.

277r CC
RO

+

RL

RP

RL+RP

112

Vacuum

Amplifiers

Tube

100-

1.

THE

RADIO

500000 ONMs

RL

2-RL= 100000OHMS
3. RL=

4. RL'

so 000 oHMs

20000

01-1M3

z30
á

u

1000

100

10000

100000

MI0-EREQUENCr GAIN

=

GMV, RL

NIGN- FREOUCNCY GAIN

a

Gm*, ¿COUPLING

e

CouT

1004000

C

FREQUENCY (C.P.S

V,.CINr2

FOR COMPROMISE HIGH REOOCNCV

Figure

XLL -

8

The variation of stage gain with frequency
in an r-c coupled pentode amplifier for various values of plate load resistance

WHERE

Amplifier

typical frequency response
curve for an R -C coupled audio
amplifier is shown in figure 8.
It is seen that the amplification is poor for the
extreme high and low frequencies. The reduced
gain at the low frequencies is caused by the
loss of voltage across the coupling capacitor.
In some cases, a low value of coupling capacitor is deliberately chosen to reduce the response of the stage to hum, or to attenuate
the lower voice frequencies for communication
purposes. For high fidelity work the product of
the grid resistor in ohms times the coupling
capacitor in microfarads should equal 25,000.
(ie.: 500,000 ohms x 0.05 µfd = 25,000).
The amplification of high frequencies falls
off because of the Miller effect of the subsequent stage, and the shunting effect of residual circuit capacities. Both of these effects
may be minimized by the use of a low value of
plate load resistor.
A

Response

The correct operating bias
for a high -mu triode such
as the GSL7, is fairly critical, and will be found to be highly variable
from tube to tube because of minute variations
in contact potential within the tube itself. A
satisfactory bias method is to use grid leak
bias, with a grid resistor of one to ten meg-

Grid Leak Bias
for High Mu Triodes

EQUALIZATION

XC AT fC

.

XC AT

f

e

CUTOFF FRCQUENC, OF AMPLIFIER

C

fC

LL I PEAKING INDUCTOR
POR

COMPROMISE LOW PRCOUENC EQUALISATION

stages is obtained by taking the product of the
voltage gains of each of the successive stages.

R -C

S

RL

Re'

Sometimes the voltage gain of an amplifier
stage is rated in decibels. Voltage gain is
converted into decibels gain through the use
of the following expression: db = 20 log
A,
where A is the voltage gain of the stage. The
total gain of cascaded voltage amplifier stages
can be obtained by adding the number of
decibels gain in each of the cascaded stages.

0

NETWORK
T C DISTRIBUTED

Ro

(Goo

vi RL)

Rs Ce °RACK
Co

C

a

=

25 TO

SO

Of0 IN PARALLEL WITH

D01 MICA

CAPACITANCE FROM AsOAC WITH 001 MICA IN PARALLEL

Figure 9
SIMPLE COMPENSATED VIDEO
AMPLIFIER CIRCUIT
Resistor RL in conjunction with coil LL
serves to flatten the high -frequency response
of the stage, while CB and R serve to equalize the low- frequency response of this simple video amplifier stage.

ohms connected directly between grid and
cathode of the tube. The cathode is grounded.
Grid current flows at all times, and the effective input resistance is about one -half the
resistance value of the grid leak. This circuit
is particularly well suited as a high gain
amplifier following low output devices, such
as crystal microphones, or dynamic micro-

phones.

resistance- capacity
coupled amplifier can
be designed to provide
a good frequency response for almost any
desired range. For instance, such an amplifier
can be built to provide a fairly uniform amplification for frequencies in the audio range of
about 100 to 20,000 cycles. Changes in the
values of coupling capacitors and load resistors can extend this frequency range to
cover the very wide range required for video
service. However, extension of the range can
only be obtained at the cost of reduced overall amplification. Thus the R -C method of
coupling allows good frequency response with
minimum distortion, but low amplification.
Phase distortion is less with R -C coupling
R -C Amplifier
General Characteristics

www.americanradiohistory.com

A

HANDBOOK

Video Frequency Amplifiers

than with other types, except direct coupling.
The R -C amplifier may exhibit tendencies to
"motorboat" or oscillate if it is used with a

high impedance plate supply.

resistance -capacitance coupling is most commonly used, there are certain circuit conditions wherein coupling methods other than
resistance capacitance are more effective.
Transformer coupling, as illustrated in figure 1013, is seldom
used at the present time between
two successive single -ended stages of an
audio amplifier. There are several reasons why
resistance coupling is favored over transformer
coupling between two successive single -ended
stages. These are: (1) a transformer having
frequency characteristics comparable with a
properly designed R -C stage is very expensive;
(2) transformers, unless they are very well
shielded, will pick up inductive hum from
nearby power and filament transformers; (3)
the phase characteristics of step -up interstage
transformers are poor, making very difficult
the inclusion of a transformer of this type
within a feedback loop; and (4) transformers
are heavy.
However, there is one circuit application
where a step-up interstage transformer is of
considerable assistance to the designer; this
is the case where it is desired to obtain a
large amount of voltage to excite the grid of a
cathode follower or of a high -power Class A
amplifier from a tube operating at a moderate
plate voltage. Under these conditions it is possible to obtain apeak voltage on the secondary
of the transformer of a value somewhat greater
than the d-c plate supply voltage of the tube
supplying the primary of the transformer.
Transformer
Coupling

Video -Frequency

6 -6

Amplifiers
A video -frequency amplifier is one which
has been designed to pass frequencies from
the lower audio range (lower limit perhaps 50
cycles) to the middle r -f range (upper limit
perhaps 4 to 6 megacycles). Such amplifiers,
in addition to passing such an extremely wide
frequency range, must be capable of amplifying this range with a minimum of amplitude,
phase, and frequency distortion. Video amplifiers are commonly used in television, pulse
communication, and radar work.

Tubes used in video amplifiers must have
high ratio of Gm to capacitance if a usable
gain per stage is to be obtained. Commonly
available tubes which have been designed for
or are suitable for use in video amplifiers are:
6AU6, 6AG5, 6AK5, 6CB6, 6AC7, 6AG7, and
6K6 -GT. Since, at the upper frequency limits
of a video amplifier the input and output
shunting capacitances of the amplifier tubes
have rather low values of reactance, low
values of coupling resistance along with
peaking coils or other special interstage coupling impedances are usually used to flatten
out the gain /frequency and hence the phase/
frequency characteristic of the amplifier.
Recommended operating conditions along with
expressions for calculation of gain and circuit
values are given in figure 9. Only a simple
two -terminal interstage coupling network is
shown in this figure.
The performance and gain -per -stage of a
video amplifier can be improved by the use
of increasingly complex two-terminal inter stage coupling networks or through the use
of four -terminal coupling networks or filters
between successive stages. The reader is referred to Terman's "Radio Engineer's Handbook" for design data on such interstage
coupling networks.
a

Push -Pull Transformer

transformer
coupling between two
stages is illustrated in
figure 10C. This interstage coupling arrangement is fairly commonly used. The system is
particularly effective when it is desired, as in
the system just described, to obtain a fairly
high voltage to excite the grids of a high power audio stage. The arrangement is also
very good when it is desired to apply feedback to the grids of the push -pull stage by
applying the feedback voltage to the lowpotential sides of the two push -pull secondaries.
Impedance coupling between two
stages is shown in figure 10D.
This circuit arrangement is seldom
used, but it offers one strong advantage over
R -C interstage coupling. This advantage is
the fact that, since the operating voltage on
the tube with the impedance in the plate circuit is the plate supply voltage, it is possible
to obtain approximately twice the peak voltage output that it is possible to obtain with
R -C coupling. This is because, as has been
Impedance

Other Interstage
Coupling Methods

Figure 10 illustrates, in addition to resistance- capacitance interstage coupling, seven
additional methods in which coupling between
two successive stages of an audio -frequency

amplifier

may

be

accomplished.

Although

Push -pull

Interstage Coupling

Coupling
6 -7

113

www.americanradiohistory.com

114

Vacuum

Tube

Amplifiers

THE

RADIO

+e

pA RESISTANCE- CAPACITANCE COUPLING

©

TRANSFORMER COUPLING

©

PUSH -PULL TRANSFORMER COUPLING

©

IMPEDANCE COUPLING

IMPEDANCE -TRANSFORMER COUPLING

0

CATHODE COUPLING

pH

+5

E©

RESISTANCE- TRANSFORMER COUPLING

+5

©

INTERSTAGE

DIR- ECT

Figure 10
COUPLING METHODS FOR AUDIO FREQUENCY VOLTAGE

mentioned before, the d -c plate voltage on an
R -C stage is approximately one -half the plate
supply voltage.
These two circuit arrangements, illustrated
former Coupling
in figures 10E and 10F,
are employed when it is
desired to use transformer coupling for the
reasons cited above, but where it is desired
that the d -c plate current of the amplifier
Impedance -Transformer
and Resistance -Trans-

+5

COUPLING

AMPLIFIERS

stage be isolated from the primary of the coupling transformer. With most types of high permeability wide -response transformers it is
necessary that there be no direct -current flow
through the windings of the transformer. The
impedance- transformer arrangement of figure
10E will give a higher voltage output from
the stage but is not often used since the plate

coupling impedance (choke) must have very
high inductance and very low distributed capacitance in order not to restrict the range of

www.americanradiohistory.com

HANDBOOK

Phase

Inverters

115

same type tubes with the values of plate voltage and load resistance to be used for the

Gw

=

- GM

G= RK GM

2G+1

RR

RP'

=

RP

=

GM

G+11

L
G

RP

G+1

=

=
=

(1+

)

CATHODE RESISTOR

GM OF EACH TUBE

Al

OF EACH TUBE

RP OF EACH TUBE

EQUIVALENT FACTORS INDICATED ABOVE BY (I) ARE
THOSE OBTAINED BY USING AN AMPLIFIER WITH A PAIR
OF SIMILAR TUBE TYPES IN CIRCUIT SHOWN ABOVE.

Figure 11
Equivalent factors for a pair of similar triodes operating as a cathode-coupled audio frequency voltage amplifier.

the transformer which

it

and

its associated

tube feed. The resistance -transformer arrange-

ordinarily quite satisfactory where it is desired to feed a transformer from a voltage amplifier stage with no
d.c.in the transformer primary.
ment of figure 10F is

The cathode coupling arrangement
of figure 10G has been widely used
only comparatively recently. One
outstanding characteristic of such a circuit is
that there is no phase reversal between the
grid and the plate circuit. All other common
types of interstage coupling are accompanied
by a 180° phase reversal between the grid
circuit and the plate circuit of the tube.
Figure 11 gives the expressions for determining the appropriate factors for an equivalent triode obtained through the use of a pair
of similar triodes connected in the cathode coupled circuit shown. With these equivalent
triode factors it is possible to use the expressions shown in figure 5 to determine the
gain of the stage at different frequencies. The
input capacitance of such a stage is less than
that of one of the triodes, the effective grid to -plate capacitance is very much less (it is
so much less that such a stage may be used
as an r -f amplifier without neutralization), and
the output capacitance is approximately equal
to the grid -to -plate capacitance of one of the
triode sections. This circuit is particularly
effective with tubes such as the 6J6, 6N7, and
6SN7 -GT which have two similar triodes in
one envelope. An appropriate value of cathode
resistor to use for such a stage is the value
which would be used for the cathode resistor
cf a conventional amplifier using one of the
Cathode

Coupling

cathode -coupled stage.
Inspection of the equations in figure 11
shows that as the cathode resistor is made
smaller, to approach zero, the Gm approaches
zero, the plate resistance approaches the Rp
of one tube, and the mu approaches zero. As
the cathode resistor is made very large the Gm
approaches one half that of a single tube of
the same type, the plate resistance approaches
twice that of one tube, and the mu approaches
the same value as one tube. But since the Gm
of each tube decreases as the cathode resistor
is made larger (since the plate current will
decrease on each tube) the optimum value of
cathode resistor will be found to be in the
vicinity of the value mentioned in the previous
paragraph.

Direct coupling between successive amplifier stages (plate
of first stage connected directly to the grid of
the succeeding stage) is complicated by the
fact that the grid of an amplifier stage must
be operated at an average negative potential with respect to the cathode of that stage.
However, if the cathode of the second amplifier stage can be operated at a potential more
positive than the plate of the preceding stage
by the amount of the grid bias on the second
amplifier stage, this direct connection between
the plate of one stage and the grid of the succeeding stage can be used. Figure 10H illustrates an application of this principle in
the coupling of a pentode amplifier stage to
the grid of a "hot- cathode" phase inverter. In
this arrangement the values of cathode, screen,
and plate resistor in the pentode stage are
chosen such that the plate of the pentode is at
approximately 0. 3 times the plate supply potential. The succeeding phase- inverter stage
then operates with conventional values of
cathode and plate resistor (same value of resistance) in its normal manner. This type of
phase inverter is described in more detail in
the section to follow.
Direct Coupling

6 -8

Phase Inverters

It is necessary in order to excite the grids
of a push -pull stage that voltages equal in
amplitude and opposite in polarity be applied
to the two grids. These voltages may be obtained through the use of a push -pull input
transformer such as is shown in figure 10C.
It is possible also, without the attendant bulk
and expense of a push -pull input transformer,
to obtain voltages of the proper polarity and

www.americanradiohistory.com

116

Vacuum Tube

Amplifiers

phase through the use of a so- called phase inverter stage. There are a large number of
phase inversion circuits which have been developed and applied Elut the three shown in
figure 12 have been found over a period of
time to be the most satisfactory from the point
of view of the number of components required
and from the standpoint of the accuracy with
which the two out -of -phase voltages are held
to the same amplitude with changes in supply
voltage and changes in tubes.
All of these vacuum tube phase inverters
are based upon the fact that a 180° phase
shift occurs within a vacuum tube between the
grid input voltage and the plate output voltage.
In certain circuits, the fact that the grid input
voltage and the voltage appearing across the
cathode bias resistor are in phase is used for
phase inversion purposes.

"Hot- Cathode"

Figure 12A illustrates the hot Phase Inverter
cathode type of phase inverter. This type of phase inverter is the simplest of the three types since
it requires only one tube and a minimum of
circuit components. It is particularly simple
when directly coupled from the plate of a
pentode amplifier stage as shown in figure
10H. The circuit does, however, possess the
following two disadvantages: (1) the cathode
of the tube must run at a potential of approximately 0.3 times the plate supply voltage
above the heater when a grounded common
heater winding is used for this tube as well
as the other heater -cathode tubes in a receiver
or amplifier: (2) the circuit actually has a
loss in voltage from its input to either of the
output grids-about 0.9 times the input voltage will be applied to each of these grids.
This does represent a voltage gain of about
1.8 in total voltage output with respect to input (grid -to -grid output voltage) but it is still
small with respect to the other two phase
inverter circuits shown.
Recommended component values for use
with a 6J5 tube in this circuit are shown in
figure 12A. If it is desired to use another tube
in this circuit, appropriate values for the operation of that tube as a conventional amplifier
can be obtained from manufacturer's tube data.
The value of RL obtained should be divided by
two, and this new value of resistance placed
in the circuit as RL. The value of Rk from
tube manual tables should then be used as
Rkl in this circuit, and then the total of Rkl
and Rk2 should be equal to RL.

"Floating Paraphase"

alternate type of
phase inverter sometimes called the "floating paraphase" is illustrated in figure 12B.
This circuit is quite often used with a 6N7
Phase Inverter

An

THE

OA

RADIO

"HOT CATHODE, PHASE INVERTER

® "FLOATING PARAPHAS6'PHASE

RL
47

INVERTER

Cc ma

RG

CC.02

22011

11

22011

G=

©

CATHODE COUPLED PHASE INVERTER

Figure 12
THREE POPULAR PHASE -INVERTER CIRCUITS WITH RECOMMENDED VALUES FOR
CIRCUIT COMPONENTS

tube, and appropriate values for the 6N7 tube
in this application are shown. The circuit
shown with the values given will give a voltage gain of approximately 21 from the input
grid to each of the grids of the succeeding
stage. It is capable of approximately 70 volts
peak output to each grid.
The circuit inherently has a small unbalance
in output voltage. This unbalance can be eliminated, if it is required for some special application, by making the resistor Rgl a few per
cent lower in resistance value than RB3.

The circuit shown in figure
12C gives approximately one half the voltage gain from the
input grid to either of the grids of the succeeding stage that would be obtained from a
single tube of the same type operating as a
conventional R -C amplifier stage. Thus, with
a 6SN7 -GT tube as shown (two 6J5's in one
Cathode -Coupled
Phase Inverter

www.americanradiohistory.com

Vacuum

HANDBOOK
.01

R5

Tube Voltmeter

117

R6

I.R
RP1

D.C.

INPUT

o

Ec

EP

_i11Figure 14
DIRECT COUPLED
D -C

Figure 13
VOLTAGE DIVIDER PHASE
INVERTER

AMPLIFIER

same amplitude as the output of

V

but of

opposite phase.
envelope) the voltage gain from the input
grid to either of the output grids will be approximately 7-the gain is, of course, 14 from
the input to both output grids. The phase
characteristics are such that the circuit is
commonly used in deriving push -pull deflection voltage for a cathode -ray tube from a
signal ended input signal.
The first half of the 6SN7 is used as an
amplifier to increase the amplitude of the applied signal to the desired level. The second
half of the 6SN7 is used as an inverter and
amplifier to produce a signal of the same
amplitude but of opposite polarity. Since the
common cathode resistor, Rk, is not by- passed
the voltage across it is the algebraic sum of
the two plate currents and has the same shape
and polarity as the voltage applied to the input grid of the first half of the 6SN7. When a
signal, e, is applied to the input circuit, the
effective grid- cathode voltage of the first
section is Ae/2, when A is the gain of the
first section. Since the grid of the second
section of the 6SN7 is grounded, the effect of
the signal voltage across Rk (equal to e/2 if
Rk is the proper value) is the same as though
a signal of the same amplitude but of opposite
polarity were applied to the grid. The output
of the second section is equal to Ae /2 if the
plate load resistors are the same for both tube
sections.

commonly used phase inverter is shown in figure 13.
The input section (V,) is connected as a conventional amplifier. The output voltage from V, is impressed on the voltage divider R, -R,. The values of R, and R,
are in such a ratio that the voltage impressed
upon the grid of V2 is 1/A times the output
voltage of V where A is the amplification
factor of V,. The output of Vt is then of the
Voltage Divider
Phase Inverter

A

D -C

6 -9

Amplifiers

Direct current amplifiers are special types
used where amplification of very slow variations in voltage, or of d -c voltages is desired.
amplifier consists of a single
A simple d -c
tube with a grid resistor across the input
terminals, and the load in the plate circuit.

A simple d -c amplifier
circuit is shown in
figure 14, wherein the
the grid of one tube is connected directly to
the plate of the preceding tube in such a
manner that voltage changes on the grid of
the first tube will be amplified by the system.
The voltage drop across the plate coupling
resistor is impressed directly upon the grid
of the second tube, which is provided with
enough negative grid bias to balance out the
excessive voltage drop across the coupling
resistor. The grid of the second tube is thus
maintained in a slightly negative position.
The d -c amplifier will provide good low frequency response, with negligible phase distortion. high frequency response is limited
by the shunting effect of the tube capacitances,
as in the normal resistance coupled amplifier.
A common fault with d -c amplifiers of all
types is static instability. Small changes in
the filament, plate, or grid voltages cannot
be distinguished from the exciting voltage.
Regulated power supplies and special balancing circuits have been devised to reduce the
effects of supply variations on these amplifiers. A successful system is to apply the
plate potential in phase to two tubes, and to
apply the exciting signal to a push -pull grid

Basic

D -C

Amplifier Circuit

www.americanradiohistory.com

118

Vacuum

Amplifiers

Tube

THE

RADIO

BALANCE
CONTROL

Figure

15

LOFTIN -WHITE
D -C AMPLIFIER

Figure

16

PUSH -PULL D -C AMPLIFIER
WITH EITHER SINGLE -ENDED
OR PUSH -PULL INPUT

circuit configuration. If the two tubes are
identical, any change in electrode voltage is
balanced out. The use of negative feedback
can also greatly reduce drift problems.
The

"Loftin -Whiter

Circuit

Two

stages

amplifier

d -c

may be arranged,

so that their plate
supplies are effectively in series, as illustrated in figure 15. This is known as a Loftin White amplifier. All plate and grid voltages
may be obtained from one master power supply
instead of separate grid and plate supplies.
A push-pull version of this amplifier (figure 16)
can be used to balance out the effects of slow
variations in the supply voltage.

6 -10

Single -ended Triode

Amplifiers
Figure 17 illustrates five circuits for the
operation of Class A triode amplifier stages.
Since the cathode current of a triode Class Al
(no grid current) amplifier stage is constant
with and without excitation, it is common
practice to operate the tube with cathode
bias. Recommended operating conditions in
regard to plate voltage, grid bias, and load
impedance for conventional triode amplifier
stages are given in the RCA Tube Manual,
RC -16.

It is possible, under certain
conditions to operate singleended triode amplifier stages
pentode and tetrode stages as well) with
excitation of sufficient amplitude that
current is taken by the tube on peaks.
type of operation is called Class A2 and

Extended Class A

Operation

(and
grid
grid

This

is characterized by increased plate -circuit
efficiency over straight Class A amplification
without grid current. The normal Class A1
amplifier power stage will operate with a plate
circuit efficiency of from 20 per cent to perhaps
35 per cent. Through the use of Class A2
operation it is possible to increase this plate
circuit efficiency to approximately 38 to 45
per cent. However, such operation requires
careful choice of the value of plate load impedance, a grid bias supply with good regulation (since the tube draws grid current on
peaks although the plate current does not
change with signal), and a driver tube with
moderate power capability to excite the grid
of the Class A2 tube.
Figures 17D and 17E illustrate two methods
of connection for such stages. Tubes such as
the 845, 849, and 304TL are suitable for such
a stage. In each case the grid bias is approximately the same as would be used for a Class
Al amplifier using the same tube, and as
mentioned before, fixed bias must be used

-

along with an audio driver of good regulation
preferably a triode stage with a 1:1 or step down driver transformer. In each case it will
be found that the correct value of plate load
impedance will be increased about 40 per cent
over the value recommended by the tube manufacturer for Class A1 operation of the tube.

Class A power amplifier
operates in such a way as
to amplify as faithfully as
possible the waveform applied to the grid of the tube. Large power output is of more importance than high voltage
amplification, consequently gain characteristics may be sacrificed in power tube design
to obtain more important power handling capabilities. Class A power tubes, such as the 45,
2A3 and 6ÁS7 are characterized by a low
amplification factor, high plate dissipation
and relatively high filament emission.
The operating characteristics of a Class A
Operation Characteristics of a Triode
Power Amplifier

www.americanradiohistory.com

A

Triode Amplifier Characteristics

HANDBOOK

E5

-(0.

119

68 x Ebb)

ll
There Ebb is the actual plate voltage of
the Class A stage, and µ is the amplification factor of the tube.
pA IMPEDANCE

3-

COUPLING

4-

5-

®

TRANSFORMER COUPLING

6-

7-

©

IMPEDANCE -TRANSFORMER COUPLING

8-

Locate the E5 bias point on the IP vs.
Ep graph where the E5 bias line crosses
the plate voltage line, as shown in figure
18. Call this point P.
Locate on the plate family of curves the
value of zero -signal plate current, I
corresponding to the operating point, P.
Locate 2 x 1, (twice the value of 1p) on
the plate current axis (Y- axis). This
point corresponds to the value of maximum signal plate current, imu.
Locate point x on the d -c bias curve at
zero volts (Eg = 0), corresponding to the
value of imax.
Draw a straight line(x - y) through points
x and P. This line is the load resistance
line. Its slope corresponds to the value
of the load resistance.
Load Resistance, (in ohms)
RL

Zsz RL

-

emu
.

¡max

- Cain
- tmin

where e is in volts, i is in amperes, and
RL is in ohms.
-e1AS

=

0 TRANSFORMER COUPLING

At OPERATION

-

AUTO
TRANSFORMER

-DIAS

©

e+e

TO

CLASS C
LOAD

CLASS Az MODULATOR WITH AUTO-TRANSFORMER COUPLING

Figure

17

Output coupling arrangements for single -ended
Class A triode audio -frequency power amplifiers.

triode amplifier employing an output transformer- coupled load may be calculated from
the plate family of curves for the particular
tube in question by employing the following
steps:
1- The load resistance should be approximately twice the plate resistance of the
tube for maximum undistorted power output. Remember this fact for a quick check

calculations.
Calculate the zero -signal bias voltage

on
2-

(Eg).

Multiply the zero - signal plate
current, Ii,, by the operating plate voltage, Ep. If the plate dissipation rating
of the tube is exceeded, it is necessary
to increase the bias (E5) on the tube so
that the plate dissipation falls within
the maximum rating of the tube. If this
step is taken, operations 2 through 8
must be repeated with the new value of

9- Check:

+e
FOR

E5.
10- For maximum power output, the peak a -c
grid voltage on the tube should swing to

2E5 on the negative cycle, and to zero bias on the positive cycle. At the peak

of the negative swing, the plate voltage reaches emu and the plate current
drops to iain On the positive swing of
the grid signal, the plate voltage drops
to envia and the plate current reaches
ima. The power output of the tube is:
Power Output (watts)
Po -

(imax

-

¡min) x (emaz

- emin)

8

where i is in amperes and

e

is in volts.

11- The second harmonic distortion generated
in a single -ended Class A triode amplifier, expressed as a percentage of the

fundamental output signal is:

www.americanradiohistory.com

120

250

MN111
to
W200 SOW.
cc

7

.

W

ai11
,50

a

RADIO

ou

f

:

THE

Amplifiers

Tube

11

ä

_

a1
11
I71
gIf/
I\

Vacuum

Figure

19

Normal single -ended pentode or beam tetrad.
audio- frequency power output stage.

xi

I.1
IMIN -

0

v

-

I

200

EMIN.

300

EG

PLATE VOLTS

400

1

EMAX.

AVERAGE PLATE CHARACTERISTICS

p. =4.2

Rp

/

/

1

100

- 2A3

OHMS
PLATE DISSIPATION =15 WATTS
=

BOO

LOAD RESISTANCE
RL

-

EMAZ
I

-

EMIN.

MAX.' IMIN.

OHMS

POWER OUTPUT
Po

(IMAX.

-IMIN) IEMAX- Ei,) WATTS
8

- IP
X

'MAX.

Figure

100 PERCENT

18

Formulas for determining the operating conditions for a Class A triode single -ended audio frequency power output stage. A typical load
line has been drawn on the average plate characteristics of a type 2A3 tube to illustrate
the procedure.

%

0. 9 Ep

2d harmonic =

(imax

- imin)

IP

iman

and the power output is somewhat less than

Ip

2

(x

Ep x Ip

100)

-imin

2

Figure 18 illustrates the above steps as
applied to a single Class A 2A3 amplifier stage.
6 -11

The operating characteristics of pentode power
amplifiers may be obtained
from the plate family of
curves, much as in the manner applied to
triode tubes. A typical family of pentode plate
curves is shown in figure 20. It can be seen
from these curves that the plate current of the
tube is relatively independent of the applied
plate voltage, but is sensitive to screen voltage. In general, the correct pentode load resistance is about
Operating Characteristics of a Pentode
Power Amplifier

SECOND HARMONIC DISTORTION
IMAX.+ IMiN.)
2

plifier stage. Tubes of this type have largely
replaced triodes in the output stage of receivers and amplifiers due to the higher plate
efficiency (30 % -40 %) with which they operate.
Tetrode and pentode tubes do, however, introduce a considerably greater amount of harmonic
distortion in their output circuit, particularly
odd harmonics. In addition, their plate circuit
impedance (which acts in an amplifier to damp
loudspeaker overshoot and ringing, and acts
in a driver stage to provide good regulation) is
many times higher than that of an equivalent
triode. The application of negative feedback
acts both to reduce distortion and to reduce
the effective plate circuit impedance of these
tubes.

Single -ended Pentode

Amplifiers
Figure 19 illustrates the conventional circuit for a single -ended tetrode or pentode am-

These formulae may be used for a quick check
on more precise calculations. To obtain the
operating parameters for Class A pentode amplifiers, the following steps are taken:
1- The imax point is chosen so as to fall on
the zero -bias curve, just above the
"knee" of the curve (point A, figure 20) .
2- A preliminary operating point, P, is determined by the intersection of the plate
voltage line, Fp, and the line of imaz /2.

www.americanradiohistory.com

Push -Pull

'HANDBOOK

Amplifiers

121

is:
Power Output (watts)

6- The power output

(¡max
.

,iAA

Po0

.

.OAD_iNE

t4/. ,OA^I¡vE

AP=PB

+ 1.41

eR

%
P(STATIC VALUE)

eMA%

is equal to:

¡max

¡max

Where IP

Figure 20

%

POWER AMPLIFIER

is the negative control grid voltage at
the operating point P

The grid voltage curve that this point
falls upon should be one that is about a
the value of ER required to cut the plate
current to a very low value (Point B).
Point B represents imin on the plate current axis (y-axis). The line ima /2 should
be located half-way between ima and

twin

trial load line is constructed about
point P and point A in such a way that
the lengths A -P and P -B are approximately equal.
hen the most satisfactory load line has
4been determined, the load resistance may
calculated:
3- A

emax

-

envia

imax

-

imin

X

RL

ER + 0. 7 F.R.

distortion is:

distortion

-

min

-

1,

2

x100
(Ix - Iy)
- ¡min
is the static plate current of
+

1.41

of the tube.

GRAPHIC DETERMINATION OF OPERATING CHARACTERISTICS OF A PENTODE

RL

2d harmonic
=

PLATE VOLTS

"V"

-4)2

Where
is the plate current at the point
on the load line where the grid voltage,
eR, is equal to: ER - 0. 7 F.R; and where
Iy is the plate current at the point where
7- The percentage harmonic

e MIN

(Ix

32

l

CHOOSE

SOrHAT

- ¡min)

5- The operating bias (ER) is the bias at

point P.

3d harmonic

(Ix

- Ty)

+ 1.41 (1x

- ly)

¡max

-imin -1.41

¡max

- tmin

6 -12

x100

Push -Pull Audio

Amplifiers
A number of advantages are obtained through
the use of the push -pull connection of two or
four tubes in an audio -frequency power amplifier. Two conventional circuits for the use
of triode and tetrode tubes in the push -pull
connection are shown in figure 21. The two
main advantages of the push -pull circuit arrangement are: (1) the magnetizing effect
of the plate currents of the output tubes is
cancelled in the windings of the output transformer; (2) even harmonics of the input signal
(second and fourth harmonics primarily) generated in the push -pull stage are cancelled
when the tubes are balanced.
The cancellation of even harmonics generated in the stage allows the tubes to be oper-

PUSH -PULL TRIODE AND TETRODE

FIGURE

distortion

21

www.americanradiohistory.com

122

Ii

e..
itdNii,
I.a
r11\I,11
Vacuum

300

Amplifiers

Tube

THE

J111111rrI

RADIO

1111NN1111NNN11NNlli' NN111111.

Ilun

.rv
IIM
i
iiii:iii:'
:
11N
/
I
.
.
1/

"um=

iGGiiGiiGiiG
rM\RJ111N.
1111N1C::NII::..Ci
'
I%
11111111NII Ci7 Ci
GGG

táiGGiiGiï

N..llrCiC
1111I
r

A
250

,

11I.1111AIi
11
,

VALUE Of
ZERO SIGNAL
PLATE CUR

N/

W/N
M
Y w1'CCl
M. :íRG

/N!!:í.
t1111P..

50

111/111/11

!!I!!1!!1!!IIlII!I!!a 111111111111111

111111111111111111,¡m1111111111N111N
11111111111111111i1111111111111111111111

1111111111111111i/1111111111.1111111111

Iiill iiiiiii%1/N1111111111111111111

1111111111111
IIIIIIN11/I
1111111111

IIIINIIII111111
i'Il:il:?'i11NN11111111111111111111111

..\!

PLATE VOLTS

11

11111

1

1111111111

Cil:i=l. 'MIN_

00

.1!
-.

II74VU'A
l....
11

o.

I.M

II

11111 P;11161111N111111111111111111111
1l:í1111D41N1 111111111111111111111111
70
e0 -30 -<0
30 -20
-0
0

-60

300

(EP)

GRID VOLTS (EG)

Figure 22
DETERMINATION OF OPERATING PARAMETERS
FOR PUSH-PULL CLASS A
TRIODE TUBES

-in

ated Class AB
other words the tubes may
be operated with bias and input signals of
such amplitude that the plate current of alternate tubes may be cut off during a portion of
the input voltage cycle. If a tube were operated
in such a manner in a single -ended amplifier
the second harmonic amplitude generated would
be prohibitively high.
Push -pull Class AB operation allows a plate
circuit efficiency of from 45 to 60 per cent to
be obtained in an amplifier stage depending
upon whether or not the exciting voltage is
of such amplitude that grid current is drawn
by the tubes. If grid current is taken on input
voltage peaks the amplifier is said to be operating Class AB2 and the plate circuit efficiency can be as high as the upper value just
mentioned. If grid current is not taken by the
stage it is said to be operating Class AB1 and
the plate circuit efficiency will be toward the
lower end of the range just quoted. In all Class
AB amplifiers the plate current will increase
from 40 to 150 per cent over the no- signal
value when full signal is applied.

The operating characteristics of push pull Class A amplifiers may also be
determined from the plate family of curves for
Operating Characteristics
of Push -Pull Class A
Triode Power Amplifier

a

particular triode tube by the following steps:
1- Erect a vertical line from the plate voltage axis (x -axis) at 0.6 Ep (figure 22),
which intersects the Eg = 0 curve. This
point of intersection (P), interpolated to
the plate current axis (y -axis) may be
taken as imp. It is assumed for simplification that imaz occurs at the point of
the zero -bias curve corresponding to
2-

0.6 Ep.
The power output obtainable from the two
tubes is:
Power output (Po)

-

i

x Ep
5

where PO is expressed in watts, imax in
amperes, and Ep is the applied plate

voltage.
3-

a preliminary load line through
point P to the Ep point located on the
x -axis (the zero plate current line). This
load line represents % of the actual plate to -plate load of the Class A tubes. Therefore:

Draw

RL

(plate -to- plate)

www.americanradiohistory.com

Ep
= 4 x

1.6 ED
max

0.6 Ep
¡Max

Class

I-ANDBOOK
where RL is expressed in ohms, Ep in
volts, and imu in amperes.
Figure 22 illustrates the above steps applied to a push -pull Class A amplifier using
two 2A3 tubes.
4- The average plate current is 0.636 Imp,
and, multiplied by the plate voltage, Ep,
will give the average watts input to the
plates of the two tubes. The power output should be subtracted from this value
to obtain the total operating plate dissipation of the two tubes. If the plate

dissipation

is excessive, a slightly

higher value of RL should be chosen to
limit the plate dissipation.
5- The correct value of operating bias, and
the static plate current for the push -pull
tubes may be determined from the Eg vs.
1p curves, which are a derivation of the
Ep vs. Ip curves for various values of
Eg.
6- The Fg vs. Ip curve may be constructed
in this manner: Values of grid bias are
read from the intersection of each grid

bias curve with the load line. These
points are transferred to the Eg vs. Ip
graph to produce a curved line, A -B. If
the grid bias curves of the Ep vs. Ip
graph were straight lines, the lines of
the Eg vs. 1p graph would also be straight
This is usually not the case. A tangent
to this curve is therefore drawn, starting
at point A', and intersecting the grid
voltage abscissa (x- axis). This intersection (C) is the operating bias point
for fixed bias operation.
7- This operating bias point may now be
plotted on the original Eg vs. 1p family
of curves (C'), and the zero-signal current produced by this bias is determined.
This operating bias point (C') does not fall
on the operating load line, as in the case of a
single -ended amplifier.
8- Under conditions of maximum power output, the exciting signal voltage swings
from zero-bias voltage to zero -bias voltage for each of the tubes on each half
of the signal cycle. Second harmonic
distortion is largely cancelled out.

6

-13

B Audio Frequency
Power Amplifiers

Class

The Class B audio- frequency power amplifier (figure 23) operates at a higher plate circuit efficiency than any of the previously
described types of audio power amplifiers.
Full- signal plate- circuit efficiencies of 60 to

B

Bt

Audio

DRIVER

Amplifiers

- BIAS

Bt

123

MOD

(GROUND FOR
ZERO WAS

OPERATING
CONDITION)

Figure 23
CLASS B AUDIO FREQUENCY
POWER AMPLIFIER

70 per cent are readily obtainable with the

tube types at present available for this type
of work. Since the plate circuit efficiency is
higher, smaller tubes of lower plate dissipation may be used in a Class B power amplifier
of a given power output than can be used in
any other conventional type of audio amplifier.
An additional factor in favor of the Class B
audio amplifier is the fact that the power input to the stage is relatively low under nosignal conditions. It is for these reasons that
this type of amplifier has largely superseded
other types in the generation of audio -frequency
levels from perhaps 100 watts on up to levels
of approximately 150,000 watts as required for
large short -wave broadcast stations.
Disadvantages of
Class B Amplifier

There are attendant dis advantageous features to the
Operation
operation of a power amplifier of this type; but all
these disadvantages can be overcome by proper
design of the circuits associated with the
power amplifier stage. These disadvantages
are: (1) The Class B audio amplifier requires
driving power in its grid circuit; this disadvantage can be overcome by the use of an
oversize power stage preceding the Class B
stage with a step -down transformer between
the driver stage and the Class -B grids. Degenerative feedback is sometimes employed
to reduce the plate impedance of the driver
stage and thus to improve the voltage regulation under the varying load presented by the
Class B grids. (2) The Class B stage requires
a constant value of average grid bias to be
supplied in spite of the fact that the grid current on the stage is zero over most of the
cycle but rises to values as high as one -third
of the peak plate current on the peak of the
exciting voltage cycle. Special regulated bias
supplies have been used for this application,
or B batteries can be used. However, a number

www.americanradiohistory.com

124

Vacuum Tube

Amplifiers

THE

of tubes especially designed for Class B audio
amplifiers have been developed which require
zero average grid bias for their operation. The
811A, 838, 805, 809, HY -5514, and TZ -40 are
examples of this type of tube. All these so-

called "zero- bias" tubes have rated operating
conditions up to moderate plate voltages
wherein they can be operated without grid
bias. As the plate voltage is increased to
to their maximum ratings, however, a small
amount of grid bias, such as could be obtained
from several 4 1/2-volt C batteries, is required.
(3), A Class B audio -frequency power amplifier or modulator requires a source of plate
supply voltage having reasonably good regulation. This requirement led to the development
of the swinging choke. The swinging choke is
essentially a conventional filter choke in
which the core air gap has been reduced. This
reduction in the air gap allows the choke to
have a much greater value of inductance with
low current values such as are encountered
with no signal or small signal being applied
to the Class B stage. With a higher value of
current such as would be taken by a Class B
stage with full signal applied the inductance
of the choke drops to a much lower value.
With a swinging choke of this type, having
adequate current rating, as the input inductor
in the filter system for a rectifier power supply, the regulation will be improved to a point
which is normally adequate for a power supply
for a Class B amplifier or modulator stage.
Calculation of Operating
Conditions of Class B
Power Amplifiers

The following procedure can be used for
the calculation of the
operating conditions
of Class B power amplifiers when they are to
operate into a resistive load such as the type
of load presented by a Class C power amplifier. This procedure will be found quite satisfactory for the application of vacuum tubes as
Class B modulators when it is desired to
operate the tubes under conditions which are
not specified in the tube operating characteristics published by the tube manufacturer. The
same procedure can be used with equal effectiveness for the calculation of the operating
conditions of beam tetrodes as Class AB2
amplifiers or modulators when the resting
plate current on the tubes (no signal condition) is less than 25 or 30 per cent of the
maximum -signal plate current.
1- With the average plate characteristics
of the tube as published by the manufacturer before you, select a point on
the Ep = E& (diode bend) line at about
twice the plate current you expect the
tubes to kick to under modulation. If
beam tetrode tubes are concerned, select

RADIO

a point at about the same amount of plate
current mentioned above, just to the
right of the region where the Ib line
takes a sharp curve downward. This will

be the first trial point, and the plate
voltage at the point chosen should be
not more than about 20 per cent of the
d -c voltage applied to the tubes if good

plate- circuit efficiency is desired.
Note down the value of ipp. and cp.,¡, at
this point.
3- Subtract the value of epm¡ from the d -c
plate voltage on the tubes.
4- Substitute the values obtained in the
following equations:
2-

P0

=

pmau(Ebb

RL_4 (Ebb

epmin)

=

Power output
from 2 tubes

emu.)

i pma:

=

Plate -to -plate load

for

2

tubes

Full signal efficiency (Nu)

78.5

Cl_evm
Ebb

/I

Effects of Speech All the above equations are
Clipping
true for sine -wave operating

conditions of the tubes concerned. However, if a speech clipper is being
used in the speech amplifier, or if it is desired
to calculate the operating conditions on the
basis of the fact that the ratio of peak power
to average power in a speech wave is approximately 4 -to-1 as contrasted to the ratio of
2 -to-1 in a sine wave-in other words, when
non- sinusoidal waves such as plain speech or
speech that has passed through a clipper are
concerned, we are no longer concerned with
average power output of the modulator as far
as its capability of modulating a Class -C amplifier is concerned; we are concerned with its
peak -power- output capability.
Under these conditions we call upon other,
more general relationships. The first of these
is: It requires a peak power output equal to
the Class -C stage input to modulate that input
fully.
The second one is: The average power output required of the modulator is equal to the
shape factor of the modulating wave multiplied by the input to the Class -C stage. The
shape factor of unclipped speech is approximately 0. 25. The shape factor of a sine wave
is 0. 5. The shape factor of a speech wave that

www.americanradiohistory.com

eoo

ï
ó

600

-111

ma.

U1

U
Ò

400

125

200

O.C.

-

o
N(Ong_.
H-201111.
agarla...
- +6a

vaLTs Ecc

p10

N'

been drawn

on the overage
characteristics of o type 811
tube.

Parameters

B

EF e 6.3 VOLTS

.u.

Figure 24
Typical Class 8 o-f amplifier
load line. The load line has

m
d'
n

NA
:Ma
,C
W! ..s/tll
Class

HANDBOOK

11iáse
-O=e=sr =
s

I

400

600

1200

_

2400

2000

1800

PLATE VOLTS (Ebb)
AVERAGE PLATE CHARACTERISTICS TYPE 811 AND 811 -A

has been passed through a clipper -filter arrangement is somewhere between 0. 25 and 0. 9
depending upon the amount of clipping that
has taken place. With 15 or 20 db of clipping
the shape factor may be as high as the figure
of 0.9 mentioned above. This means that the
audio power output of the modulator will be
90% of the input to the Class -C stage. Thus
with a kilowatt input we would be putting
900 watts of audio into the Class -C stage for
100 per cent modulation as contrasted to perhaps 250 watts for unclipped speech modulation of 100 per cettt.

Figure 24 shows a set of
plate characteristics for
a type 811A tube with a
load line for Class B operation. Figure 25
lists a sample calculation for determining the
proper operating conditions for obtaining approximately 185 watts output from a pair of
the tubes with 1000 volts d -c plate potential.
Also shown in figure 25 is the method of determining the proper ratio for the modulation
transformer to couple between the 811's or
811A's and the anticipated final amplifier
which is to operate at 2000 plate volts and
175 ma. plate current.
Sample Calculation
for 811A Tubes

Lion shown in figure 25. or by reference to the
published characteristics on the tubes to be
used. (2) Determine the load impedance which
will be presented by the Class C amplifier
stage to be modulated by dividing the operating
plate voltage on that stage by the operating
value of plate current in amperes. (3) Divide
the Class C load impedance determined in (2)

SAMPLE CALCULATION
CONDITION:

2 TYPE 811 TUBES, Ebb, = 1000
INPUT TO FINAL STAGE, 350 W.
PEAR POWER OUTPUT NEEDED. 350 IS% = 370
FINAL AMPLIFIER Ebb = 2000 V.
FINAL AMPLIFIER Ib = .175 A.
FINAL AMPLIFIER ZL = -22SISL = 11400 R
.175

EXAMPLE

CHOSE POINT ON 811

TO RIGHT OF

IP

MAX.

IG MAX.

PEAK PO

Ebb'

F /G.

EP MIN.

A.

EG MAX. _

RL

=

4 X

NP

=

78.5 (1

.410

_

=

X

900

-

)

1

(.9)

76.5

=

WO (AVERAGE WITH SINE WAVE)

WIN

=

- 260

Ió.5

Ib (MAXIMUM
WO PEAR

=

=

369

W.

=

X80

=

=

70.5 "b
POIPEAR)_I813W

W.

WITH SINE WAVE)

100

DRIVING POWER

80

8800 n.

=

:9000

24 )

+100

A.

.100

.410 0 (1000 -10o)

=

CHARACTERISTICS JUST

Ecc. (PO /NT X.

=.410
_

W.

=

=

260 MA

e W

WZ PR

-

W.

TRANSFORMER:

The method illustrated
in figure 25 can be used
in general for the determination of the proper transformer ratio to
couple between the modulator tube and the
amplifier to be modulated. The procedure can
be stated as follows: (1) Determine the proper
plate -to-plate load impedance for the modulator
tubes either by the use of the type of calculaModulation Transformer

Calculation

114

ZP

sew

TURNS RATIO

=

- 1.29

LA
ZP

=

1

29

=

1.14 STEP UP

Figure 25
Typical calculation of operating conditions for
a Class B a -f power amplifier using a pair of
type 811 or 811A tubes. Plate characteristics
and load line shown in figure 24.

www.americanradiohistory.com

126

Vacuum

Tube

Amplifiers

above by the plate -to -plate load impedance for
the modulator tubes determined in (1) above.
The ratio determined in this way is the sec ondary-to- primary impedance ratio. (4) Take
the square root of this ratio to determine the
secondary-to- primary turns ratio. If the turns
ratio is greater than one the use of a step -up
transformer is required. If the turns ratio as
determined in this way is less than one a stepdown transformer is called for.
If the procedure shown in figure '25 has
been used to calculate the operating conditions
for the modulator tubes, the transformer ratio
calculation can be checked in the following
manner: Divide the plate voltage on the modulated amplifier by the total voltage swing on
the modulator tubes: 2 (Ebb
e, 0). This ratio
should be quite close numerically to the transformer turns ratio as previously determined.
The reason for this condition is that the ratio
between the total primary voltage and the d-c
plate supply voltage on the modulated stage
is equal to the turns ratio of the transformer,
since a peak secondary voltage equal to the
plate voltage on the modulated stage is required to modulate this stage 100 per cent.

-

Use of Clipper Speech

Amplifier with Tetrode
Modulator Tubes

clipper speech
amplifier is used in
conjunction with a Class

current.

As stated previously, a
Class B audio amplifier
requires the driving stage
to supply well -regulated audio power to the
grid circuit of the Class B stage. Since the
performance of a Class B modulator may easily
be impaired by an improperly designed driver
stage, it is well to study the problems incurred in the design of the driver stage.
The grid circuit of a Class B modulator may
be compared to a variable resistance which
decreases in value as the exciting grid voltage is increased. This variable resistance appears across the secondary terminals of the
driver transformer so that the driver stage is
Class

B

Modulators

RADIO

called upon to deliver power to a varying load.
For best operation of the Class 13 stage, the
grid excitation voltage should not drop as the
power taken by the grid circuit increases.
These opposing conditions call for a high order of voltage regulation in the driver stage
plate circuit. In order to enhance the voltage
regulation of this circuit, the driver tubes must
have low plate resistance, the driver transformer must have as large a step -down ratio
as possible, and the d-c resistance of both
primary and secondary windings of the driver
transformer should be low.
The driver transformer should reflect into
the plate circuit of the driver tubes a load of
such value that the required driving power is
just developed with full excitation applied to
the driver grid circuit. If this is done, the
driver transformer will have as high a stepdown ratio as is consistent with the maximum
drive requirements of the Class B stage. If
the step -down ratio of the driver transformer is
too large, the driver plate load will be so
high that the power required to drive the Class
B stage to full output cannot be developed.
If the step-down ratio is too small the regulation of the driver stage will be impaired.

When a

B modulator stage, the
plate current on that stage will kick to a
higher value with modulation(due to the greater
average power output and input) but the plate
dissipation on the tubes will ordinarily be
less than with sine -wave modulation. However,
when tetrode tubes are used as modulators,
the screen dissipation will be much greater
than with sine -wave modulation. Care must
be taken to insure that the screen dissipation rating on the modulator tubes is not exceeded under full modulation conditions with
a clipper speech amplifier. The screen dissipation is equal to screen voltage times screen

Practical Aspects of

THE

Driver Stage
Calculations

The parameters for the driver
stage may be calculated from
the plate characteristic curve, a
sample of which is shown in figure 24. The
required positive grid voltage (eg -m,$) for the
811A tubes used in the sample calculation is
found at point X, the intersection of the load
line and the peak plate current as found on the
y -axis.

This is + 80 volts. If a vertical line
is dropped from point X to intersect the dotted

current curves, it will be found that the
current for a single 811A at this value of
voltage is 100 milliamperes (point Y).
peak grid driving power is therefore
80 x 0.100 = 8 watts. The approximate average
driving power is 4 watts. This is an approximate figure because the grid impedance is not
constant over the entire audio cycle.
A pair of 2A3 tubes will be used as drivers,
operating Class A, with the maximum excitation to the drivers occuring just below the
point of grid current flow in the 2A3 tubes.
The driver plate voltage is 300 volts, and the
grid bias is -62 volts. The peak power developed in the primary winding of the driver
transformer is:
grid
grid
grid
The

Peak Power (Pe) = 2R1 I

PE&

,a

`Rp + RIwhere it is the amplification factor of the
driver tubes (4.2 for 2A3). Eg is the peak grid
swing of the driver stage (62 volts). Rp is the
(watts)

www.americanradiohistory.com

Cathode Follower Amplifier

HANDBOOK
plate resistance of one driver tube (800 ohms).
RL is % the plate -to -plate load of the driver
stage, and Pp is 8 watts.
Solving the above equation for RL, we
obtain a value of 14,500 ohms load, plate -toplate for the 2A3 driver tubes.
The peak primary voltage is:
epri = 2RL x

g

Ft,

+RL

493 volts

and the turns ratio of the driver transformer
(primary to % secondary) is:
epri

-=
493

=

eg(ma:)

80

6.15:1

Plate Circuit One of the commonest
distortion in a Class
Impedance

causes of
B

modu-

lator is incorrect load impedance
in the plate circuit. The purpose
of the Class B modulation transformer is to
take the power developed by the modulator
(which has a certain operating impedance) and
transform it to the operating impedance imposed by the modulated amplifier stage.
If the transformer in question has the same
number of turns on the primary winding as it
has on the secondary winding, the turns ratio
is 1:1, and the impedance ratio is also 1:1. If
a 10,000 ohm resistor is placed across the
secondary terminals of the transformer, a reflected load of 10,000 ohms would appear
across the primary terminals. If the resistor
is changed to one of 2376 ohms, the reflected
primary impedance would also be 2376 ohms.
If the transformer has twice as many turns
on the secondary as on the primary, the turns
ratio is 2:1. The impedance ratio is the square
of the turns ratio, or 4:1. If a 10,000 ohm
resistor is now placed across the secondary
winding, a reflected load of 2,500 ohms will
appear across the primary winding.
Matching

It can be seen

from the
above paragraphs that the
Class B modulator plate
load is entirely dependent upon the load
placed upon the secondary terminals of the
Class B modulation transformer. If the secondary load is incorrect, certain changes will
take place in the operation of the Class B
modulator stage.
When the modulator load impedance is too
low, the efficiency of the Class B stage
is reduced and the plate dissipation of the
tubes is increased. Peak plate current of
the modulator stage is increased, and saturation of the modulation transformer core may
result. "Talk -back" of the modulation trans-

Effects of Plate
Circuit Mis -match

127

former may result if the plate load impedance
of the modulator stage is too low.
When the modulator load impedance is too
high, the maximum power capability of the

stage is reduced. An attempt to increase the
output by increasing grid excitation to the
stage will result in peak -clipping of the audio
wave. In addition, high peak voltages may be
built up in the plate circuit that may damage
the modulation transformer.

6 -14

Cathode- Follower
Power Amplifiers

The cathode -follower is essentially a power
output stage in which the exciting signal is
applied between grid and ground. The plate is
maintained at ground potential with respect to
input and output signals, and the output signal
is taken between cathode and ground.

Figure 26 illustrates four
types of cathode - follower
power amplifiers in common usage and figure 27 shows the output
impedance (Ro), and stage gain (A) of both
triode and pentode(or tetrode) cathode- follower
stages. It will be seen by inspection of the
equations that the stage voltage gain is always
less than one, that the output impedance of
the stage is much less than the same stage
operated as a conventional cathode -return
amplifier. The output impedance for conventional tubes will be somewhere between
100 and 1000 ohms, depending primarily on
the transconductance of the tube.
This reduction in gain and output impedance for the cathode -follower comes about
since the stage operates as though it has 100
per cent degenerative feedback applied between
its output and input circuit. Even though the
voltage gain of the stage is reduced to a value
less than one by the action of the degenerative
feedback, the power gain of the stage (if it is
operating Class A) is not reduced. Although
more voltage is required to excite a cathode follower amplifier than appears across the load
circuit, since the cathode "follows" along
with the grid, the relative grid-to- cathode voltage is essentially the same as in a conventional amplifier.
Types of Cathode-

Follower Amplifiers

Although the cathode -follower gives no voltage
gain, it is an effective
power amplifier where it is desired to feed a
low- impedance load, or where it is desired to
feed a load of varying impedance with a signal
having good regulation. This latter capability
Use of Cathode-

Follower Amplifiers

www.americanradiohistory.com

128

Vacuum

Tube

Amplifiers

THE
TRIODE

-U

ucr
J,1

+1

Re (CATHODE

PENTODE:

Ro(cAr.,00E

A

=

RADIO

A

+
GM

RL

L

+Rp

(Rn,+Rea)
RK,

Rao

RL
)

RL(.U+I

Ri

+Rn2+ RL'

R

1+RL Gu

G.. Rea

Figure 27
Equivalent

factors for pentode (or tetrad.)
cathode- follower power amplifiers.

plifier tube, the components

Figure 26
CATHODE-FOLLOWER OUTPUT
CIRCUITS FOR AUDIO OR
VIDEO AMPLIFIERS

makes the cathode follower particularly effective as a driver for the grids of a Class B
modulator stage.
The circuit of figure 26A is the type of amplifier, either single -ended or push -pull, which
may be used as a driver for a Class B modulator or which may be used for other applications such as feeding a loudspeaker where unusually good damping of the speaker is desired. If the d -c resistance of the primary of
the transformer T2 is approximately the correct
value for the cathode bias resistor for the am-

Rk and Ck need
not be used. Figure 26B shows an arrangement
which may be used to feed directly a value of
load impedance which is equal to or higher
than the cathode impedance of the amplifier
tube. The value of Cc must be quite high,
somewhat higher than would be used in a conventional circuit, if the frequency response of
the circuit when operating into a low- impedance load is to be preserved.
Figures 26C and 26D show cathode -follower
circuits for use with tetrode or pentode tubes.
Figure 26C is a circuit similar to that shown
in 26A and essentially the same comments
apply in regard to the components Rk and Ck
and the primary resistance of the transformer
T2. Notice also that the screen of the tube is
maintained at the same signal potential as the
cathode by means of coupling capacitor Cd.
This capacitance should be large enough so
that at the lowest frequency it is desired to
pass through the stage its reactance will be
low with respect to the dynamic screen -tocathode resistance in parallel with Rd T2 in
this stage as well as in the circuit of figure
26A should have the proper turns (or impedance) ratio to give the desired step -down or
step -up from the cathode circuit to the load.
Figure 26D is an arrangement frequently used
in video systems for feeding a coaxial cable of
relatively low impedance from a vacuum -tube
amplifier. A pentode or tetrode tube with a
cathode imped*tce as a cathode follower
(1 /G,a) approximately the same as the cable
impedance should be chosen. The 6AG7 and
6AC7 have cathode impedances of the same
order as the surge impedances of certain types
of low- capacitance coaxial cable. An arrangement such as 26D is also usable for feeding
coaxial cable with audio or r -f energy where
it is desired to transmit the output signal
over moderate distances. The resistor Rk is
added to the circuit as shown if the cathode
impedance of the tube used is lower than the

www.americanradiohistory.com

HANDBOOK

Feedback

characteristic impedance of the cable. If the
output impedance of the stage is higher than
the cable impedance a resistance of appropriate value is sometimes placed in parallel
with the input end of the cable. The values
of Cd and Rd should be chosen with the same
considerations in mind as mentioned in the
discussion of the circuit of figure 26C above.

INPUT SIGNAL ES

The
may

6 -15

Feedback Amplifiers

It is possible to modify the characteristics
of an amplifier by feeding back a portion of
the output to the input. All components, circuits and tubes included between the point
where the feedback is taken off and the point
where the feedback energy is inserted are
said to be included within the feedback loop.
An amplifier containing a feedback loop is
said to be a feedback amplifier. One stage or
any number of stages may be included within
the feedback loop. However, the difficulty of
obtaining proper operation of a feedback amplifier increases with the bandwidth of the
amplifier, and with the number of stages and
circuit elements included within the feedback
loop.

The gain and phase
shift of any amplifier
are functions of frequency. For any amplifier containing a feedback loop to be completely stable the gain of
such an amplifier, as measured from the input
back to the point where the feedback circuit
connects to the input, must be less than one
Gain and Phase -shift
in Feedback Amplifiers

uTPUT

E

A

VOLTAGE AMPLIFICATION WITH FEEDBACK
1

cathode follower
conveniently be

used asa method of coupling r -f or i -f energy between two units separated a considerable distance. In such an
application a coaxial cable should be used to
carry the r -f or i -f energy. One such application would be for carrying the output of a v -f -o
to a transmitter located a considerable distance from the operating position. Another
application would be where it is desired to
feed a single -sideband demodulator, an FM
adaptor, or another accessory with intermediate frequency signal from a communications receiver. A tube such as a 6CB6 connected in a manner such as is shown in figure
26D would be adequate for the i -f amplifier
coupler, while a 6L6 or a 6AG7 could be used
in the output stage of a v -f -o as a cathode
follower to feed the coaxial line which carries
the v -f-o signal from the control unit to the
transmitter proper.

AMPLIFIER
GAIN= A

FEEDBACK OR B PATH

A

The Cathode -Follower
in R -F Stages

129

Amplifiers

FEEDBACK IN DECIBELS

=

GAIN IN ABSENCE

=

B

8

-A
OF

B
FEEDBACK

FRACTION OF OUTPUT VOLTAGE FED BACK

=

NEGATIVE FOR NEGATIVE FEEDBACK

IS

20 LOG

(1

-A8)

- 20 L0G MID FRED- GAIN WITHOUT FEEDBACK
MIDFREO. GAIN WITH FEEDBACK

DISTORTION WITH FEEDBACK

RD

_

DISTORTION WITHOUT FEEDBACK
(1
8)

-A

RN

_
1

-Aa

(1+

-)

WHERE

RD =OUTPUT IMPEDANCE OF AMPLIFIER WITH FEEDBACK

RNA OUTPUT IMPEDANCE
RL

=

OF

AMPLIFIER WITHOUT FEEDBACK

LOAD IMPEDANCE INTO WHICH AMPLIFIER OPERATES

Figure 28
FEEDBACK AMPLIFIER RELATIONSHIPS

at the frequency where the feedback voltage
is in phase with the input voltage of the amplifier. If the gain is equal to or more than
one at the frequency where the feedback voltage is in phase with the input the amplifier
will oscillate. This fact imposes a limitation
upon the amount of feedback which may be
employed in an amplifier which is to remain
stable. If the reader is desirous of designing
amplifiers in which a large amount of feedback is to be employed he is referred to a
book on the subject by H. W. Bode.

Feedback may be either negative
positive, and the feedback voltage may be proportional either to
output voltage or output current. The most
commonly used type of feedback with a -f or
video amplifiers is negative feedback proportional to output voltage. Figure 28 gives
the general operating conditions for feedback
amplifiers. Note that the reduction in distortion is proportional to the reduction in gain of
the amplifier, also that the reduction in the
output impedance of the amplifier is somewhat
greater than the reduction in the gain by an
amount which is a function of the ratio of the
Types of
Feedback

or

H. W. Bode,

fier

Dsign,'

and Feedback AmpliVan Nostrand Co., 250 Fourth Ave.,

"Network Analysis
D.

New York 3, N. Y.

www.americanradiohistory.com

130

Vacuum

THE

Amplifiers

Tube

RADIO

Figure 29 illustrates a very simple and effective application of negative voltage feedback to an output pentode or tetrode amplifier
stage. The reduction in hum and distortion
may amount to 15 to 20 db. The reduction in
the effective plate impedance of the stage will
be by a factor of 20 to 100 dependent upon the
operating conditions. The circuit is commonly
used in commercial equipment with tubes such
as the 6SJ7 for VI and the 6V6 or 6L6 for V2.
O° FEEDBACK

20 LOG

e

a

20L04

GAIN OP BOTH sTNGES

R2

* R. RZ(G..V2

Ra

+RR

I

¡l
I

=

RO)

(VOLTAGE CAIN

[

Goo,

(

6 -16

OrV2))

R2

)

::.!-:1)
RN

;

R(G..vz Ro)

111

WHERE.

RNR°
RD

=

R2

OUTPUT

R, %RD
R, +R2
Rz
GN.z Ro
RCrLECTt0 LOAD IMPEDANCE

.NEDPNCE

RN

=

-

RN

ON

V2

(USUALLY ABOUT S00 R)

PEED°ACN RESISTOR

R2

iRZ+RN(GwV2RO))(.+

PLATE IMPEDANCE

R

)

Vacuum -Tube Voltmeters

The vacuum -tube voltmeter may be considered
to be a vacuum -tube detector in which the
rectified d -c current is used as an indication
of the magnitude of the applied alternating
voltage. The vacuum tube voltmeter (v.t.v.m.)
consumes little or no power and it may be
calibrated at 60 cycles and used at audio or
radio frequencies with little change in the

calibration.

or V2

si mple v.t.v.m. is
shown in figure 30.
The plate load may be
a mechanical device, such as a relay or a
meter, or the output voltage may be developed
across a resistor and used for various control purposes. The tube is biased by Ec and
a fixed value of plate current flows, causing
a fixed voltage drop across the plate load
resistor, Rp. When a positive d -c voltage is
applied to the input terminals it cancels part
of the negative grid bias, making the grid
more positive with respect to the cathode.
This grid voltage change permits a greater
amount of plate current to flow, and develops
a greater voltage drop across the plate load
resistor. A negative input voltage would decrease the plate current and decrease the
voltage drop across Rp, The varying voltage
drop across Rp may be employed as a control
voltage for relays or other devices. When it is
desired to measure various voltages, a voltage
Basic

Figure 29
SHUNT FEEDBACK CIRCUIT
FOR PENTODES OR TETRODES
This circuit requires only the addition of
one resistor, R2, to the normal circuit for
such an application. The plate impedance
and distortion Introduced by the output
stage are materially reduced.

output impedance of the amplifier without
feedback to the load impedance. The reduction
in noise and hum in

those

stages included

within the feedback loop is proportional to the
reduction in gain. However, due to the reduction in gain of the output section of the amplifier somewhat increased gain is required of
the stages preceding the stages included within the feedback loop. Therefore the noise and
hum output of the entire amplifier may or may
not be reduced dependent upon the relative
contributions of the first part and the latter
part of the amplifier to hum and noise. If most
of the noise and hum is coming from the stages
included within the feedback loop the undesired signals will be reduced in the output
from the complete amplifier. It is most frequently true in conventional amplifiers that
the hum and distortion come from the latter
stages, hence these will be reduced by feedback, but thermal agitation and microphonic
noise come from the first stage and wilt not
be reduced but may be increased by feedback
unless the feedback loop includes the first
stage of the amplifier.

D -C Vacuum Tube Voltmeter

A

Figure 30
SIMPLE VACUUM TUBE
VOLTMETER

www.americanradiohistory.com

Vacuum Tube Voltmeters

HANDBOOK

131

ZERO- ADJUST

Figure 31

D -C

VACUUM TUBE VOLTMETER

Figure 32

BRIDGE -TYPE VACUUM TUBE

VOLTMETER
range switch (figure 31) may precede the v.t.
v.m. The voltage to be measured is applied to
voltage divider, R1, R2, R3, by means of the
"voltage range" switch. Resistor R4 is used
to protect the meter from excessive input
voltage to the v.t.v.m. In the plate circuit of
the tube an additional battery and a variable
resistor ( "zero adjustment ") are used to
balance out the meter reading of the normal
plate current of the tube. The zero adjustment

potentiometer can be so adjusted that the
meter M reads zero current with no input voltage to the v.t.v.m. When a d -c input voltage
is applied to the circuit, current flows through
the meter, and the meter reading is proportional to the applied d -c voltage.
The Bridge -type

V.T.V.M.

Another important use
of a d-c amplifier is to
show the exact point of

voltages. This is
done by means of a bridge circuit with two
d -c amplifiers serving as two legs of the
bridge (figure 32). With no input signal, and
with matched triodes, no current will be read
on meter M, since the IR drops across Ri and
R2 are identical. When a signal is applied to
one tube, the IR drops in the plate circuits
become unbalanced, and meter M indicates
the unbalance. In the same way, two d -c voltages may be compared if they are applied to
the two input circuits. When the voltages are
equal, the bridge is balanced and no current
flows through the meter. If one voltage changes,
the bridge becomes unbalanced and indication
of this will be noted by a reading of the meter.
balance between two

d -c

For the purpose of analysis,
the operation of a modern
v.t.v.m. will be described. The lleatbkit V -7A
isa fit instrument for such adescription, since
it is able to measure positive or negative d -c
potentials, a -c r -m -s values, peak -to-peak
values, and resistance. The circuit of this
unit is shown in figure 33. A sensitive 200 d -c

A Modern VTVM

microammeter is placed in the cathode circuit
of a 12AU7 twin triode. The zero adjust control
sets up a balance between the twosections of
the triode such that with zero input voltage
applied to the first grid, the voltage drop across
each portion of the zero adjust control is the
same. Under this condition of balance the
meter will read zero. When a voltage is applied
to the first grid, the balance in the cathode
circuits is upset and the meter indicates the
degree of unbalance. The relationship between
the applied voltage on the first grid and the
meter current is linear and therefore the meter
can be calibrated with a linear scale. Since
the tube is limited in the amount of current
it can draw, the meter movement is elec-

tronically protected.
The maximum test voltage applied to the
12AU7 tube is about 3 volts. Higher applied
voltages are reduced by a voltage divider
which has a total resistance of about 10
megohms. An additional resistance of 1- megohm
is located in the d -c test prod, thereby permitting measurements to be made in high impedance circuits with minimum disturbance.
The rectifier portion of the v.t.v.m. is shown
in figure 34. When a -c measurements are desired, a 6AL5 double diode is used as a full
wave rectifier to provide a d -c voltage pro portionalto the applied a-c voltage. This d -c

voltage is applied through the voltage divider
string to the 12AU7 tube causing the meter to
indicate in the manner previously described.
The a -c voltage scales of the meter are calibrated in both RMS and peak-to -peak values.
In the 1.5, 5, 15, 50, and 150 volt positions
of the range switch, the full a -c voltage being
measured is applied to the input of the 6AL5
full wave rectifier. On the 500 and 1500 volt
positions of the range switch, a divider network reduces the applied voltage in order to
limit the voltage input to the 6AL5 to a safe
recommended level.

www.americanradiohistory.com

132

THE

HEATHKIT PEAKTO -PEAK
MODEL V -7A

Figure

VTVM

33

www.americanradiohistory.com

RADIO

Vacuum Tube Voltmeters

HANDBOOK

02
A -C

150v

133

SNIELDED PROBE CASE

6AL5

INPUT 0---I

22

OM5

MAX

MEG

ff

VTVM
INPUT
JACK

TO
DC

COAA,L LINE

A 7

MEG

OOSi

=PROBE

TIP

Ce-70 S

Figure

34

Figure

FULL -WAVE RECTIFIER
FOR V.T.V.M.

35

-F PROBE SUITABLE
FOR USE IN IKC -100 MC
R

RANGE

The a -c calibrate control (figure 33) is used
to obtain the proper meter deflection for the
applied a -c voltage. Vacuum tubes develop a
contact potential between tube elements. Such
contact potential developed in the diode would
cause a slight voltage to be present at all
times. This voltage is cancelled out by proper
application of a bucking voltage. The amount
of bucking voltage is controlled by the a -c
balance control. This eliminates zero shift
of the meter when switching from a -c to d -c

readings.
For resistance measurements, a 1.5 volt
battery is connected through a string of multipliers and the external resistance to be measured, thus forming a voltage divider across
the battery, and a resultant portion of the
battery voltage is applied to the 12AU7 twin

triode. The meter scale is calibrated in resistance (ohms) for this function.
Test Probes

Auxiliary test probes may
be used with the v.t.v.m.
to extend the operating range, or to measure
radio frequencies with high accuracy. Shown
in figure 35 is a radio frequency probe which
provides linear response to over 100 megacycles. A crystal diode is used as a rectifier,
and d-c isolation is provided by a .005 uufd
capacitor. The components of the detector are
mounted within a shield at the end of a length
of coaxial line, which terminates in the d-c
input jack of the v.t.v.m. The readings obtained are RM1S, and should be multiplied by
1.414 to convert to peak readings.

www.americanradiohistory.com

CHAPTER SEVEN

High

Fidelity Techniques

The art and science of the reproduction of
sound has steadily advanced, following the
major audio developments of the last decade.
Public acceptance of home music reproduction
on a "high fidelity" basis probably dates from
the summer of 1948 when the Columbia L -P
microgroove recording techniques were introduced.
The term high fidelity refers to the reproduction of sound in which the different distortions of the electronic system are held below
limits which are audible to the majority of
listeners. The actual determination, therefore,
of the degree of fidelity of a music system is
largely psychological as it is dependent upon
the ear and temperament of the listener. By and
large, a rough area of agreement exists as to
what boundaries establish a "hi -fi" system. To
enumerate these boundaries it is first necessary
to examine sound itself.

As shown in figure 1, the sound wave of
the fork has frequency, period, and pitch. The
frequency is a measure of the number of vibrations per second of the sound. A fork tuned
to produce 261 vibrations per second is tuned
to the musical note of middle -C. It is of interest to note that any object vibrating, moving,
or alternating 261 times per second will produce a sound having the pitch of middle -C.
The pitch of a sound is that property which is
determined by the frequency of vibration of
the source, and not by the source itself. Thus
an electric dynamo producing 261 c.p.s. will
have a hum -pitch of middle -C, as will a siren,
a gasoline engine, or other object having the
same period of oscillation.

))111I)

7 -1 The Nature of Sound

))'

III

Experiments with a simple tuning fork in
the seventeenth century led to the discovery
that sound consists of a series of condensations
and rarefactions of the air brought about by
movement of air molecules. The vibrations of
the prongs of the fork are communicated to
the surrounding air, which in turn transmits
the agitation to the ear drums, with the result
that we hear a sound. The vibrating fork produces a sound of extreme regularity, and this
regularity is the essence of music, as opposed to
noise which has no such regularity.

TUNING FORK

Figure

1

VIBRATION OF TUNING FORK PRODUCES A SERIES OF CONDENSATIONS
AND RAREFACTIONS OF AIR MOLECULES. THE DISPLACEMENT OF AIR
MOLECULES CHANGES CONTINUALLY
WITH RESPECT TO TIME, CREATING
A SINE WAVE OF MOTION OF THE
DENSITY VARIATIONS.

134

www.americanradiohistory.com

Nature of Sound
FREQUENCY (CYCLES
NOTE

D

C

EQUAL-

TEMPERED 261.0

E

F

PER SECOND)

G

A

B

-I

W

I

H

C'

293.7 329.6 349.2 392.0 440.0 493.9 523.21

SCALE

I

Figure 2
EQUAL- TEMPERED SCALE

CONTAINS TWELVE INTERVALS, EACH OF
WHICH IS 1.06 TIMES THE FREQUENCY OF THE NEXT LOWEST. THE HALFINCLUDE
THE
INTERVALS
TONE
ABOVE NOTES PLUS FIVE ADDITIONAL NOTES: 277.2, 311.1, 370, 415.3,
466.2 REPRESENTED BY THE BLACK
KEYS OF THE PIANO.
THE

135

C

7

T/ME

-J

a.

2

Figure 3
THE COMPLEX SOUND OF A MUSICAL
INSTRUMENT IS A COMBINATION OF
SIMPLE SINE -WAVE SOUNDS, CALLED
HARMONICS. THE SOUND OF LOWEST
FREQUENCY IS TERMED THE FUNDAMENTAL. THE COMPLEX VIBRATION
OF A CLARINET REED PRODUCES A
SOUND SUCH AS SHOWN ABOVE.

The Musical

The musical scale is composed
of notes or sounds of various
frequencies that bear a pleasing
aural relationship to one another. Certain combinations of notes are harmonious to the ear
if their frequencies can be expressed by the
simple ratios of 1:2, 2:3, 3:4, and 4:5. Notes
differing by a ratio of 1:2 are said to be separated by an octave.
The frequency interval represented by an
octave is divided into smaller intervals, forming the musical scales. Many types of scales
have been proposed and used, but the scale of
the piano has dominated western music for the
last hundred or so years. Adapted by J. S.
Bach, the equal- tempered scale ( figure 2) has
twelve notes, each differing from the next by
the ratio 1:1.06. The reference frequency, or
American Standard Pitch is A, or 440.0 cycles.
Scale

Harmonics and
Overtones

The complex sounds pro duced by a violin or a wind
instrument bear little resemblance to the simple sound wave of the tuning
fork. A note of a clarinet, for example (when
viewed on an oscilloscope) resembles figure 3.
Vocal sounds are even more complex than this.
In 1805 Joseph Fourier advanced his monumental theorem that made possible a mathematical analysis of all musical sounds by showing that even the most complex sounds are
made up of fundamental vibrations plus harmonics, or overtones. The tonal qualities of
any musical note may be expressed in terms of
the amplitude and phase relationship between
the overtones of the note.
To produce overtones, the sound source must
be vibrating in a complex manner, such as is
shown in figure 3. The resulting vibration is
a combination of simple vibrations, producing
a rich tone having fundamental, the octave
tone, and the higher overtones. Any sound

-

-

no matter how complex
can be analyzed
into pure tones, and can be reproduced by a
group of sources of pure tones. The number
and degree of the various harmonics of a tone
and their phase relationship determine the

quality of the tone.
For reproduction of the highest quality, these
overtones must be faithfully reproduced. A musical note of 523 cycles may be rich in twentieth order overtones. To reproduce the original quality of the note, the audio system must
be capable of passing overtone frequencies of
the order of 11,000 cycles. Notes of higher
fundamental frequency demand that the audio
system be capable of good reproduction up to
the maximum response limit of the human
ear, in the region of 15,000 cycles.
Reproduction

Many factors enter into the
problem of high quality audio
reproduction. Most important
of these factors influence the overall design of
the music system. These are:
Restricted frequency range.
Nonlinear distortions.
Limitations

1-

23-Transient distortion.
4- Nonlinear frequency response.
-Phase distortion.
6- Noise, "wow ", and "flutter ".
5

A restricted frequency range of reproduction
will tend to make the music sound "tinny"
and unrealistic. The fundamental frequency
range covered by the various musical instruments and the human voice lies between 15
cycles and 9,000 cycles. Overtones of the instruments and the voice extend the upper
audible limit of the music range to 15,000
cycles or so. In order to fully reproduce the
musical tones falling within this range of frequencies the music system must be capable of
flawlessly reproducing all frequencies within
the range without discrimination.

www.americanradiohistory.com

THE RADIO

High Fidelity Techniques

136

BASIC LIMITS FOR HIGH FIDELITY AND
"GOOD QUALITY+ REPRODUCTION
TYPE OF
DISTORTION

RESTRICTED
FREQUENCY
RANGE

LIMIT
HIGH FIDELITY

20 -IS000 CPS

INTER MODULATION
DISTORTION AT
FULL OUTPUT

4

%

2

%

HARMONIC
DISTORTION AT
FULL OUTPUT

WOW"
HUM AND NOISE

"GOODY REPRODUCTI I

SO

-10000

10 W.

S

0.1%
-70

DB BELOW

FULL OUTPUT

CPS

%

1W.

-50

OB BELOW

FULL OUTPUT

Figure 4

Nonlinear qualities such as harmonic and
intermodulation (IM) distortion are extremely
objectionable and are created when the output
of the music system is not exactly proportional
to the input signal. Nonlinearity of any part
of the system produces spurious harmonic frequencies, which in turn lead to unwanted beats
and resonances. The combination of harmonic
frequencies and intermodulation products produce discordant tones which are disagreeable
to the ears.
The degree of intermodulation may be measured by applying two tones f1 and f: of known
amplitude to the input of the amplifier under
test. The relative amplitude of the difference
tone (f2-f1) is considered a measure of the
intermodulation distortion. Values of the order
of 4% IM or less define a high fidelity music
amplifier.
Response of the music system to rapid transient changes is extremely important. Transient peaks cause overloading and shock- excitation of resonant circuits, leaving a "hang- over"
effect that masks the clarity of the sound. A
system having poor transient response will not
sound natural to the ear, even though the distortion factors are acceptably low.
Linear frequency response and good power
handling capability over the complete audio
range go hand in hand. The response should
be smooth, with no humps or dips in the curve
over the entire frequency range. This requirement is particularly important in the electromechanical components of the music system,
such as the phonograph pickup and the loudspeaker.
Phase distortion is the change of phase angle between the fundamental and harmonic
frequencies of a complex tone. The output

wave envelope therefore is different from the
envelope of the input wave. In general, phase
distortion is difficult to hear in sounds having
complex waveforms and may be considered to
be sufficiently low in value if the IM figure of
the amplifier is acceptable.
Noise and distortions introduced into the
program material by the music system must
be kept to a minimum as they are particularly
noticeable. Record scratch, turntable "rumble",
and "flutter" can mar an otherwise high quality system. Inexpensive phonograph motors do
not run at constant speed, and the slight variations in speed impart a variation in pitch
( wow) to the music which can easily be heard.
Vibration of the motor may be detected by the
pickup arm, superimposing a low frequency
rumble on the music.
The various distortions that appear in a
music system are summarized in figure 4, together with suggested limits within which the
system may truly be termed "high fidelity."

7 -2 The Phonograph
The modern phonograph record is a thin
disc made of vinylite or shellac material. Disc
rotation speeds of 78.3, 33 1/3, and 45 r.p.m.
are in use, with the older 78.3 r.p.m. speed
gradually being replaced by the lower speeds.
A speed of 16 2/3 r.p.m. is used for special
"talking book" recordings. A continuous groove
is cut in the record by the stylus of the recording machine, spiralling inward towards the
center of the record. Amplitude variations in
this groove proportional to the sound being
recorded constitute the means of placing the
intelligence upon the surface of the record.
The old 78.3 r.p.m. recordings were cut approximately 100 grooves per inch, while the
newer "micro- groove" recordings are cut approximately 250 grooves per inch. Care must
be taken to see that the amplitude excursions
of one groove do not fall into the adjoining
groove. The groove excursions may be controlled by the system of recording, and by
equalization of the recording equipment.
The early commercial phonograph records were cut with a
mechanical- acoustic system that
produced a constant velocity characteristic with
the amplitude of cut increasing as the recorded
frequency decreased ( figure 5A) . When the
recording technique became advanced enough
to reproduce low audio frequencies, it was
necessary to reduce the amplitude of the lower
frequencies to prevent overcutting the record.
A crossover point near 500 cycles was chosen,
Recording
Techniques

www.americanradiohistory.com

HANDBOOK

High Fidelity Amplifier

CONSTANT AMPLITUDE,

137

CROSSOVER
FREQUENCY
CONSTANT
VELOCITY

SURFACE NO /SE

SURFACE NO /SE
L__ 4

f

4

SO

100

200

-1-

1

4

500

1000

FREQUENCY

2000

5000

50

100

I

500

FREQUENCY

(cvs)

1000

\

2000

5000
'

(CPS)

CONSTANT AMPLITUDE BELOW CROSSOVER
FREQUENCY. CONSTANT VELOCITY ABOVE
CROSSOVER FREQUENCY

CONSTANT VELOCITY RECORDING

V--

t

200

CROSSOVER

FREQUENCY

SURFACE NO

SURFACE NOISE
50

100

200

1000

500

2000

5000

SO

200

100

500

FREQUENCY

FREQUENCY (,P-)
CONSTANT AMPLITUDE BELOW CROSSOVER
FREQUENCY, NIGH FREQUENCY PRE -EMPHASIS
ABOVE CROSSOVER FREQUENCY

1000

/SE

2000

solo°

(cps)

RESPONSE OF RECORD OF SC PLAYED ON
PROPERLY COMPENSATED EQUIPMENT

Figure 5
MODERN PHONOGRAPH RECORD EMPLOYS CONSTANT AMPLITUDE CUT
BELOW CROSSOVER POINT AND HIGH
FREQUENCY PRE- EMPHASIS (BOOST)
ABOVE CROSSOVER FREQUENCY.
"RIAA" PLAYBACK

10

20

40

70100

300 500

CURVE

1000

3000

10000 30000

FREQUENCY. CPS

The most popular types
of pickup cartridges in
use today are the high
impedance crystal unit, and the low impedance variable reluctance cartridge. The
crystal pickup consists of a Rochelle salt
element which is warped by the action of
the phonograph needle, producing an electrical impulse whose frequency and amplitude are proportional to the modulation of
the record groove. One of the new "transducer"
crystal cartridges is shown in figure 6. When
working into a high impedance load, the output of a high quality crystal pickup is of the
The Phonograph
Pickup

and a constant amplitude groove was cut below
this frequency ( figure 5B). This system does
not reproduce the higher audio notes, since the
recording level rapidly drops into the surface
noise level of the record as the cutting frequency is raised. The modern record employs
pre-emphasis of the higher frequencies to boost
them out of the noise level of the record
(figure 5C) . When such a record is played
back on properly compensated equipment, the
audio level will remain well above the background noise level, as shown in figure 5D.

www.americanradiohistory.com

138

THE RADIO

High Fidelity Techniques

Figure 8

Figure 6
NEW CRYSTAL "TRANSDUCER"
CARTRIDGE PROVIDES HIGH FIDELITY OUTPUT AT RELATIVELY
HIGH LEVEL

order of one -half volt or so. Inexpensive crystal
units used in 78 r.p.m. record changers and
ac -dc phonographs may have as much as two
or three volts peak output. The frequency response of a typical high quality crystal pickup
is shown in figure 7.
The variable reluctance pickup is shown in
figure 8. The reluctance of the air gap in a
magnetic circuit is changed by the movement
of the phonograph needle, creating a variable
voltage in a small coil coupled to the magnetic
lines of force of the circuit. The output impedance of the reluctance cartridge is of the
order of a few hundred ohms, and the output
is approximately 10 millivolts.
For optimum performance, an equalized preamplifier stage is usually employed with the
reluctance pickup. The circuit of a suitable unit
is shown in figure 9. Equalization is provided
by R5, R,, and C., with a low frequency crossover at about 500 cycles. Total equalization is
15 db. High frequency response may be limited
by reducing the value of R. to 5,000
15,000
ohms.
The standard
records has a tip
the microgroove
has a tip radius

-

pickup stylus for 78 r.p.m.
radius of .0025 inch, whereas
(33 1/3 and 45 r.p.m.) stylus
of .001 inch. Many pickups,

"RELUCTANCE" CARTRIDGE
STANDARD PICK -UP FOR

IS

MUSIC SYSTEM.

Low stylus pressure of four grams insures
minimum record wear. Dual stylus is used

having two needle tip diameters for long
playing and 78 R.P.M. recordings.

therefore, are designed to have interchangeable
cartridges or needles to accomodate the different groove widths.

7 -3

The High Fidelity Amplifier

A block diagram of a typical high fidelity
system is shown in figure 10. A preamplifier
is used to boost the output level of the phonograph pickup, and to permit adjustment of input selection, volume, record compensation,
and tone control. The preamplifier may be
mounted directly at the phonograph turntable
position, permitting the larger power amplifier
to be placed in an out of the way position.
The power amplifier is designed to operate
from an input signal of a volt or so derived
from the preamplifier, and to build this signal
to the desired power level with a minimum
amount of distortion. Maximum power output
levels of ten to twenty watts are common for
home music systems.
The power supply provides the smoothed,
d -c voltages necessary for operation of the preamplifier and power amplifier, and also the

6SC7
SNORT

a

w

OUTPUT

LEADS TO
RELUCTANCE
CARTRIDGE

+10

Rts

+5Rz

33M

Z 0
a -5
20

r

e

R3

50

too

Re
200

500

FREQUENCY

1000 2000 5000

68K

10000

C4 C5

T-

e

Rs
33K

e+ 100 v

HIGH PHONOGRAPH
QUALITY
CRYSTAL
CARTRIDGE. (ELECTROVOICE
56 -DS
POWER POINT TRANSDUCER)
RESPONSE

,33K

(cps)

Figure 7
FREQUENCY

Ce
.0

RT

66

=

TC3
_ .01

-10
Lai

¢

66K

4

OF

Figure 9
PREAMPLIFIER SUITABLE FOR USE WITH
LOW LEVEL RELUCTANCE CARTRIDGE.

www.americanradiohistory.com

i

HANDBOOK
!PHONOGRAPH

High Fidelity Amplifier

139

+20

RI

f

1

M

SPEAKER
ENCLOSURE

POWER
SUPPLY

-20

20

100

SO

200

SOO

FREQUENCY

I.

Figure 10
BLOCK DIAGRAM OF HIGH FIDELITY
MUSIC SYSTEM.

1000 2000

N

TREBLE

5000 10000

(CeS)

Figure 12
FREQUENCY RESPONSE CURVES FOR
THE BASS AND TREBLE BOOST
AND ATTENUATION CIRCUITS
OF FIGURE 11.

TREBLE BOOST
AND ATTENUATION

BASS BOOST
AND ATTENUATION

1

SPONSE
LOUD
SPEAKER

I

v

-t

I

POWER
PREAMPLIFIER+ AMPLIFIER

S

TREBLE BOOST

BASS BOOST

TWEETERI

INPUT

RI

eóosr

C,

Rz

80ó$r
OUTPUT

OUTPUT

ATTENUATE

ArrENUATE

C2

Rn

EQUIVALENT CIRCUITS

EQUIVALENT CIRCUITS

ATTENUATE

ATTENUATE

BOOST

IN.

Ri
OUT.

OUT.

C2

Rz

SIMPLE

Equalizer

networks are employed in high fidelity equipment to 1)- tailor the response curve of the system to obtain the correct
compensate
overall frequency response, 2)
for inherent faults in the program material,
merely to satisfy the hearing preference
3
of the listener. The usual compensation networks are combinations of RC and RL networks that provide a gradual attenuation over
a given frequency range. The basic RC networks suitable for equlizer service are shown
in figure 11. Shunt capacitance is employed
for high frequency attenuation, and series
capacitance is used for low frequency attenuation. A combination of these simple a -c
voltage dividers may be used to provide almost
any response, as shown in figure 12. It is
common practice to place equalizers between
two vacuum tubes in the low level stages of
the preamplifier, as shown in figure 13. Bass
and treble boost and attenuation of the order

Tone

Compensaton

-to

C1

our

hold its own in the race for true fidelity.
Speaker efficiency runs from about 10% for
cone units to nearly 40% for high frequency
tweeters. The frequency response of any speaker is a function of the design and construction
of the speaker enclosure or cabinet that mounts
the reproducer.

O

Figure 11
CIRCUITS MAY BE

R -C

USED FOR BASS AND TREBLE
BOOST OR ATTENUATION.

filament voltages (usually a -c) for the heaters
of the various amplifier tubes.
The loudspeaker is a device which couples
the electrical energy of the high fidelity system to the human ear and usually limits the
overall fidelity of the complete system. Great
advances in speaker design have been made in
the past years, permitting the loudspeaker to

)-or

+1zAx7

.04

E. OUTPUT

220

INPUT

Figure 13

AND TREBLE LEVEL
CONTROLS, AS
EMPLOYED IN THE
HEATHKIT WA -P2
PREAMPLIFIER.

BASS

B+
BASS CONTROL

www.americanradiohistory.com

TREBLE CONTROL

K

140

THE RADIO

High Fidelity Techniques
2N190

2N190

IRK

2N 190

12V

1K

VARIABLE
RELUCTANCE

PICKUP

e

3 9K

OUTPUT

or;
TREBLE

BASS

Figure 14
TRANSISTORIZED HIGH- FIDELITY PREAMPLIFIER FOR USE WITH
RELUCTANCE PHONOGRAPH CARTRIDGE.

f100-

Loudness
Compensaron

PAIN L EVEL

+90
+BO

+70

+60
+50

+40
+30

LOWER

LIMIT

OF

HEARING

+20
+10
0

10

20

50

100

200

500

FREQUENCY

1d00 2000 5000

10000

(CPS)

200

Figure 15
THE "FLETCHER -MUNSON" CURVE
ILLUSTRATING THE INTENSITY
RESPONSE OF THE HUMAN EAR.

of 15 db may be obtained from such a circuit.

simple transistorized preamplifier using
this type of equalizing network is shown in
figure 14.
A

The minimum threshold of
hearing and the maximum
threshold of pain vary greatly with the frequency of the sound as shown
in the Fletcher- Munson curves of figure 15.
To maintain a reasonable constant tonal balance as the intensity of the sound is changed
it is necessary to employ extra bass and treble
boost as the program level is decreased. A
simple variable loudness control is shown in
figure 16 which may be substituted for the
ordinary volume control used in most audio
equipment.
The Power

The power amplifier stage of
the music system must supply
driving power for the loudspeaker. Commercially available loudspeakers
are low impedance devices which present a
Amplifier

mVIO

o
w +5

z

o

a

O1

0

SPEAKER

-5

RESPONSE

ß:

R -10
w
aw
o")

V

14
'

O

-RESONANT FREQUENCY OF
-SPEAKER - BAFFLE COMBINATION

a

a
m_
iiI
Illi

20

RI

-R2 -R

31 THREE SECTION

POTENTIOMETER, /RC
THE FOLLOWING,

TYPE, BUILT OF
R1 - lAC P011 -135

R2 -/RC MOLT /SECT /ON M13 -137
R3 - /RC MOLT /SECT /ON M13-128

10

20

50

100

SPEAKER

200

500

FREQUENCY

Figure 16
VARIABLE LOUDNESS CONTROL FOR
USE IN LOW IMPEDANCE PLATE
CIRCUITS. MAY BE PURCHASED
AS IRC TYPE LC -1 LOUDNESS
CONTROL.

1000

2000 5000

4

10000

(CPS)

Figure 17
IMPEDANCE AND FREQUENCY

RESPONSE OF "4 -OHM" 12 -INCH
SPEAKER PROPERLY MOUNTED IN
MATCHING BAFFLE.

www.americanradiohistory.com

Ir

20000

HANDBOOK

High Fidelity Amplifier

141

FEEDBACK RESISTOR

VJ'

Figure 18

TYPICAL TRIODE
AMPLIFIER WITH
FEEDBACK LOOP.

*

° MATCHED

varying load of two to nearly one hundred
ohms to the output stage ( figure 17) . It is
necessary to employ a high quality output
transformer to match the loudspeaker load to
the relatively high impedance plate circuit of
the power amplifier stage. In general, push pull amplifiers are employed for the output
stage since they have even harmonic cancelling
properties and permit better low frequency
response of the output transformer since there
is no d -c core saturation effect present.
To further reduce the harmonic distortion
and intermodulation inherent in the amplifier
system a negative feedback loop is placed
around one or more stages of the unit. Frequency response is thereby improved, and the
output impedance of the amplifier is sharply
reduced, providing a very low source impedance for the loudspeaker.

PAIR RESISTORS

Shown in figure 18 is a basic push -pull
triode amplifier, using inverse feedback around
the power output and driver stage. A simple
triode inverter is used to provide 180- degree
phase reversal to drive the grid circuit of the
power amplifier stage. Maximum undistorted
power output of this amplifier is about 8 watts.
A modification of the basic triode amplifier
is the popular Williamson circuit (figure 19)
developed in England in 1947. This circuit
rapidly became the "standard of comparison"
in a few short years. Pentode power tubes are
connected as triodes for the output stage, and
negative feedback is taken from the secondary
of the output transformer to the cathode of
the input stage. Only the most linear portion
of the tube characteristic curve is used. Although that portion has been extended by
higher than normal plate supply voltage, it

FEEDBACK RESISTOR

SK-ISK

6SN7GT

65N7GT

807

0.25

INPUT

OUTPUT

22K
NJ

30KK{

= 1

10 20

4-400V.
AT 140 MA

Figure 19

U. S. VERSION OF BRITISH

OUTPUT AT

LESS

THAN

"WILLIAMSON" AMPLIFIER PROVIDES 10 WATTS POWER
2 °o

INTERMODULATION DISTORTION. 6SN7 STAGE
DIRECT COUPLING.

www.americanradiohistory.com

USES

142

THE RADIO

High Fidelity Techniques
807/5881

TO

FEEDBACK

CIRCUIT
0

25

FROM
05N7GT
PHASE

OUTPUT

INVERTER

0.25

807/5881

NOTE, P/N CONNECT IONS ARE
POR 807 TUBES

f00V

Figure 20

"ULTRA- LINEAR" CONFIGURATION OF WILLIAMSON AMPLIFIER DOUBLES POWER OUTPUT, AND REDUCES IM LEVEL. SCREEN TAPS ON OUTPUT TRANSFORMER PERMIT
"SEMI -TETRODE" OPERATION.

is only a fraction of the curve normally used
in amplifiers. Thus a comparatively low output
power level is obtained with tubes capable of

much more efficient operation under less
stringent requirements. With 400 volts applied to the output stage, a power output of 10
watts may be obained wtih less than 2% inter modulation distortion.
A recent variation of the Williamson circuit involves the use of a tapped output transformer. The screen grids of the push -pull amplifier stage are connected to the primary taps,
allowing operating efficiency to approach that
of the true pentode. Power output in excess
of 25 watts at less than 2% intermodulation dis-

Figure 21

"BABY HI -FI" AMPLIFIER IS DWARFED
BY 12 -INCH SPEAKER ENCLOSURE

This miniature music system is capable of excellent performance in the small home or
apartment. Preamplifier, bass and treble controls, and volume control are all incorporated

in the unit. Amplifier provides 4 watts output
at 4 IM distortion.

tortion may be obtained with this circuit
(figure 20) .

7 -4

Amplifier Construction

Wiring

Assembly and layout of high
fidelity audio amplifiers follows the general technique described for other forms of electronic equipment.
Extra care, however, must be taken to insure
that the hum level of the amplifier is extremely low. A good hi -fi system has excellent response in the 60 cycle region, and even a
minute quantity of induced a -c voltage will be
disagreeably audible in the loudspeaker. Spurious eddy currents produced in the chassis by
the power transformer are usually responsible
for input stage hum.
To insure the lowest hum level, the power
transformer should be of the "upright" type
instead of the "half-shell" type which can
couple minute voltages from the windings to
a steel chassis. In addition, part of the windings
of the half -shell type project below the chassis
where they are exposed to the input wiring of
the amplifier. The core of the power transformer should be placed at right angles to the
core of a nearby audio transformer to reduce
spurious coupling between the two units to a
Techniques

minimum.
It is common practice in amplifier design
to employ a ground bus return system for all
audio tubes. All grounds are returned to a
single heavy bus wire, which in turn is
grounded at one point to the metal chassis.
This ground point is usually at the input jack
of the amplifier. When this system is used,
a -c chassis currents are not coupled into the
amplifying stages. This type of construction is
illustrated in the amplifiers described later in
this chapter.

www.americanradiohistory.com

HANDBOOK

Amplifier Construction
NO
FOR

RI=e.2

220

.05

PHONO INPUT

J,
K^

e n SPEAKER

6AQ5
12AU7

12AÚ7

143

470K
.05

e

Sp

MMr

W

R110
1,T2

len
SPKR

(HIGH LEVEL)
100

K

M

3

1.e

1.9K

SISOK

R

10011

OS

M

70 K

BASE

5

6AQ5

VOLUME
CONTROL
CONTROL

1,7

TREBLE
CONTROL

+203

10
450V

V.

1211
I

+210V
6X5

TI

CH1
3K 2W
2

115V
ti

IxÇ,A
=4,5

9

12AÚ7

T,- 260 -0-260

?

124Ú7

.20C111

=3

TOC1C
3

4

NOTES
ALL RESISTORS 0.3 WATT
UNLESS OTHERWISE SPECIFIED
2. ALL CAPACITOR VALUES IN MF
UNLESS OTHERWISE SPECIFIED
3. RESISTORS MARKED A ARE
MATCHED PAIRS
1.

=

4
6.405

RAO5

Figure 22
SCHEMATIC, "BABY HI -FI" AMPLIFIER
CH,-1.5 henry at 200 ma. Chicago Standard
6.3 volts at 4.0

volts at 90 ma.,
amp., upright mounting. Chicago -Standard
PC -8420.
T,-10 K, CT. to 8, 16 ohms. Peerless (Altec)
S -510F.

Care should be taken to reduce the capacitance to the chassis of high impedance circuits,
or the high frequency response of the unit will
suffer. Shielded "bath -tub" type capacitors
should not be used for interstage coupling capacitors. Tubular paper capacitors are satisfactory. These should be spaced well away from
the chassis.
It is a poor idea to employ the chassis as a
common filament return, especially for low
level audio stages. The filament center-tap of
the power transformer should be grounded,
and twisted filament wires run to each tube
socket. High impedance audio components and
wiring should be kept clear of the filament
lines, which may even be shielded in the vicinity of the input stage. In some instances, the
filament center tap may be taken from the arm
of a low resistance, wirewound potentiometer
placed across the filament pins of the input
tube socket. The arm of this potentiometer is
grounded, and the setting of the control is adjusted for minimum speaker hum.

7 -5 The "Baby Hi

Fi"

A definite need exists for a compact, high
fidelity audio amplifier suitable for use in the
small home or apartment. Listening tests have
shown that an average power level of less than

C,A- B-

C- 30 -20 -10

C -2327.
Aid. 350 volt. Mallory Fp -330.7

NOTE -Feedback loop returns to 8 ohm tap on T,
when 8 ohm speaker is used.

one watt in a high efficiency speaker will provide a comfortable listening level for a small
room, and levels in excess of two or three watts
are uncomfortably loud to the ear. The "Baby
Hi -Fi" amplifier has been designed for use
in the small home, and will provide excellent
quality at a level high enough to rattle the
windows.
Designed around the new Electro -Voice miniature ceramic cartridge, the amplifier will provide over 4 watts power, measured at the secondary of the output transformer. At this level,
the distortion figure is below 1 %, and the IM
figure is 4 %. At normal listening levels, the IM
is much lower, as shown in figure 24.
The Amplifier

The schematic of the ampli fier is shown in figure 22.
Bass and treble boost controls are incorporated in the circuit, as is the
volume control. A dual purpose 12AU7 double
triode serves as a voltage amplifier with cathode degeneration. A simple voltage divider
network is used in the grid circuit to prevent
amplifier overloading when the ceramic cartridge is used. The required input signal for
maximum output is of the order of 0.3 volts.
The output level of the Electro -Voice cartridge
is approximately twice this, as shown in figure
7. The use of the high -level cartridge eliminCircuit

www.americanradiohistory.com

144

THE RADIO

High Fidelity Techniques

Figure 23
UNDER -CHASSIS

VIEW OF

"BABY HI -FI"

Low level audio stages are

at

upper left, with components
mounted between socket pins
and potentiometer controls.
6X5 socket is at lower cen-

ter of

photo

with

filter

choke CH, at right. Feedback resistor R, is at left
of rectifier socket.

ates the necessity of high gain amplifiers required when low level magnetic pickup heads
are used. Problems of hum and distortion introduced by these extra stages are thereby

eliminated, greatly simplifying the amplifier.
The second section of the 12AU7 is used for
bass and treble boost. Simple R -C networks
are placed in the grid circuit permitting gain
boost of over 12 db at the extremities of the
response range of the amplifier.
A second 12AU7 is employed as a direct
coupled "hot-cathode" phase inverter, capacitively coupled to two 6AQ5 pentode connected

output tubes. The feedback loop is run from
the secondary of the output transformer to the
cathode of the input section of the phase inverter.
The power supply of the "Baby Hi -Fi" consists of a 6X5 -GT rectifier and a capacitor input filter. A second R -C filter section is used
to smooth the d -c voltage applied to the
12AU7 tubes. A cathode-type rectifier is used
in preference to the usual filament type to prevent voltage surges during the warm -up period
of the other cathode -type tubes.
Amplifier

The complete amplifier is
built upon a small "amplifier
foundation" chassis and cover
measuring 5 "x7 "x6" (Bud CA- 1754). Height
of the amplifier including dust cover is 6 ".
The power transformer (T,) and output transformer (Tl) are placed in the rear corners of
the chassis, with the'6X5 -GT rectifier socket
placed between them. The small filter choke
(CH,) is mounted to the wall of the chassis
and may be seen in the under -chassis photograph of figure 23. The four audio tubes are
placed in a row across the front of the chassis.
Viewed from the front, the 12AU7 tubes are
to the left, and the 6AQ5 tubes are to the right.
The three section filter capacitor (C,A, B, C)
is a chassis mounting unit, and is placed between the rectifier tube and the four audio
tubes. Since the chassis is painted, it is important that good grounding points be made at
each tube socket. The paint is cleared away
Construction

s-

t

o

a

3

EQUIVALENT SINE WAVE WATTS

Figure 24

INTERMODULATION CURVE FOR
"BABY HI -Fl" AS MEASURED ON
HEATHKIT INTERMODULATION
ANALYZER.

www.americanradiohistory.com

HANDBOOK

"Baby Hi -Fi"

145

Figure 25

TYPICAL
INTERMODULATION
TEST OF AUDIO
AMPLIFIER.

Audio tones of two frequencies are applied to input of amplifier under testi
and amplitude of "sum" or
"difference" frequency is
measured, providing relative
inter -modulation figure.

beneath the socket bolt heads, and lock nuts
are used beneath the socket retaining nuts to
insure a good ground connection. All ground
leads of the first 12AU7 tube are returned to
the socket, whereas all grounds for the rest of
the circuit are returned to a ground lug of

filter capacitor G.
Since the input level to the amplifier is of
the order of one-half volt, the problem of
chassis ground currents and hum is not so
prevalent, as is the case with a high gain input
stage.

Phonograph -type coaxial receptacles are
mounted on the rear apron of the chassis, serving as the input and output connections. The
four panel controls (bass boost, treble boost,
volume, and a -c on) are spaced equidistant
across the front of the chassis.
Amplifier
Wiring

The filament wiring should be
done first. The center -tap of the
filament winding is grounded to
a lug of the 6X5 -GT socket ring, and the 6.3
volt leads from the transformer are attached
to pins 2 and 7 of the same socket. A twisted
pair of wires run from the rectifier socket to
the right -hand 6AQ5 socket (figure 23). The
filament leads then proceed to the next 6AQ5
socket and then to the two 12AU7 sockets in
turn.
The 12AU7 preamplifier stage is wired next.
A two terminal phenolic tie -point strip is
mounted to the rear of the chassis, holding the
12K decoupling resistor and the positive lead

of the 10 µfd., 450 -volt filter capacitor. All
B -plus leads are run to this point. Most of the
components of the bass and treble boost system
may be mounted between the tube socket terminals and the terminals of the two potentiometers. The feedback resistor R, is mounted
between the terminal of the coaxial output
connector and a phenolic tie -point strip placed
beneath an adjacent socket bolt.
When the wiring has been completed and
checked, the amplifier should be turned on,
and the various voltages compared with the
values given on the schematic. It is important
that the polarity of the feedback loop is correct.
The easiest way to reverse the feedback polarity
is to cross -connect the two plate leads of the
6AQ5 tubes. If the feedback polarization is
incorrect, the amplifier will oscillate at a supersonic frequency and the reproduced signal will
sound fuzzy to the ear. The correct connection
may be determined with the aid of an oscilloscope, as the oscillation will be easily found.
The builder might experiment with different
values of feedback resistor RI, especially if a
speaker of different impedance is employed.
Increasing the value of R1 will decrease the
degree of feedback. For an 8 -ohm speaker, Rt
should be decreased in value to maintain the
same amount of feedback.

This amplifier was used in conjunction with
General Electric S-1201A 12 -inch speaker
mounted in an Electro -Voice KD6 Aristocrat
speaker enclosure which was constructed from
a

www.americanradiohistory.com

High Fidelity Techniques

THE RADIO

kit. The reproduction was extremely smooth,
with good balance of bass and treble.

requirements and the absence of expensive
power and audio transformers it is more economical than conventional amplifiers of similar
performance.

146
a

7 -6

A Transformerless
25 Watt Music

Amplifier

The Amplifier

The output stage of this unusual amplifier is the single
ended, push -pull type as
shown in figure 27. The quiescent current is
equal in both tubes with no d.c. current flowing through the speaker load. The absence of
an output transformer allows 40 db of feedback to be app'ied by connecting the voice
coil of the speaker directly to the cathode of
the 12ÁT7 phase- inverter driver. In addition
to its distortion reducing characteristic, the
application of feedback serves to reduce the
hum voltage which might otherwise be present.
As the gain within the feedback loop is essentially unity, an additional voltage amplifier
is used (with separate feedback) to build the
input voltage up to the voice coil level.
The power supply is a double half-wave selenium rectifier circuit developing +140 volts
and -140 volts with respect to ground. The
supply uses large filter capacitors, and no
Circuit

Because the output transformer is usually
the weakest link in both frequency response
and power output of an audio amplifier,
several methods have been used to drive loudspeakers directly from the output tubes. These
have either used non -conventional high -impedance loudspeakers, have been very inefficient,
or have had low power output capabilities.
The amplifier described in this section
drives a conventional 16 ohm loudspeaker
with normal class A amplifier efficiency, and
supplies 25 watts of low distortion output
throughout the audio range (figure 26). The
amplifier requires an input signal of approximately one volt to drive it to maximum output. The unit attains its high performance
through the use of 40 db of inverse feedback.
Because of the relatively simple power supply

Figure 26.
25 -WATT

TRANS FORMERLESS
AMPLIFIER PROVIDES ULTIMATE IN
LISTENING
PLEASURE FOR THE
"GOLDEN EAR."
Amplifier employs three
triode tubes in
single-ended push-pull
configuration for maximum fidelity. The output
6082

tubes are placed across
right end of chassis.
65N7 phase inverter is at
rear, center; and low
level stages are at the
front, center. To the left
are the power supply filter capacitors. In this

particular amplifier, the
40 ohm, 20 watt filadropping resistor
and a
iron core reactor
was used in its place
(left, rear corner of the
chassis). Across front of
chassis are (I. to r.):
power switch, input lack,
and output stage balancing potentiometer.
ment
was

small

www.americanradiohistory.com

eliminated

HANDBOOK

Transformerless Amplifier

12 AT 7

L

I
a

147

2 A T7

40

±T
+

40
150

+ 150

loo

150

47K

K

INPUT

K

6082

6SN7 -GT
0.1

*

0.1

6082

;ff-6
+

Sl

6082

56 K

5

5M

56K
4-

40

+ 150

00

100

2

6K*

5

8K

1611
OUTPUT

2

680

40
150

. IM

5

I

1.5M

M

M

1.8

10K

K

1K

4

1M

39K

00

100

56K

10
K

4

4

4

1.2
M

+250

VOLTS

+,o VOLTS
9

4,5

65147

4062 6082 6062

12AT7

6

7

7

6

6

7

11Hf--`
SR 1.5Kz

40
S6W

9

i

7

560

2.5 K
10

SI

115V1.,

NOTE:
1.

O%

40/150

12 AT7

r

n

ONE SIDE OP

70 CHASSIS.

LINE

A.

roo

fOWW

W

o

*140

óI

SR2 +aoo

GROUNDED

of

S RS

SR 4

0.5

160

ï5S

-140

VOLTS

10

A
A1°

%

T

%

REACTANCE
NETWORKS

2. RESISTORS MARRED R ARE
MATCHED PA /RS.

4.SRI.

Qx

P300500

ALL RESISTORS I -WATT UNLESS
OTHERWISE NOTED.

3. CAPACITOR VALUES

VOLTS

GIVEN IN //FD.

75 MA., I50 VOLT.

5.SRz, SR3 =500MA.,

I I

15

150 VOLT.

150

SR4

8.29

10K

27

K

12K

Figure 27.
SCHEMATIC OF 25 -WATT MUSIC AMPLIFIER

extra filtering is required. The output impedance of the supply is extremely low. To obtain higher voltage for the low level stages,
additional selenium rectifiers are used in a
voltage- adding configuration to obtain +250
and -250 volts.

about -70 volts, but for this class of service
the bias is held at -60 volts. A bias control
is provided for one set of tubes so that the
d.c. current flowing in the tubes may be
equalized, and to insure that no d.c. current
flows through the speaker voice coil.

Circuit Details

The type 6082 tube is not rated for use
with fixed bias unless a limiting resistor is
added in either the plate or the cathode circuit. Although this circuit does not use such
resistors, their omission is feasible only because the tubes are used under quiescent conditions well below maximum ratings. With
tubes of this type, it may be expected that the
average current through the voice coil will
drift with time but the presence of this un-

The complete schematic of
this amplifier is given in
figure 27. Three type 6082 double triodes are
employed in the output stage. These are 26.5
volt versions of the popular 6AS7G. These
tubes are capable of 700 milliamperes of peak
plate current per triode section at the plate
voltage employed. The choice of the 6082 is
an economy measure to allow the use of a
series heater string. These tubes cut off at

www.americanradiohistory.com

High Fidelity Techniques

THE RADIO

balance current will generally be of little
concern. In any event, the circuit has been
designed so that the output stage can be conveniently rebalanced.

voltage than the upper group.
In the first voltage amplifier, bias is obtained from unbypassed cathode resistors since
the loss of gain can easily be tolerated. The
phase inverter- driver, however, has fixed bias
applied to the grid from the -140 volt
supply, since maximum gain is desired within
the main feedback loop.

148

The Voltage

The low level stages are all
operated Class A with conventional circuitry. A separate
driver is needed for each side of the output
circuit, as insufficient output is obtained from
the phase inverter to drive the output tubes
directly. One side of the phase inverter has a
larger load than the other, since the input to
the lower group of output tubes has the
speaker impedance in the cathode. This causes
degeneration and necessitates higher input
Amplifier

Figure 28.
UNDER -CHASSIS VIEW OF
TRANSFORMERLESS AMPLIFIER
Output tube sockets are at left, with power
supply components at right. Components of

preamplifier stages are grouped about the
center sockets, mounted between socket pins
and phenolic tie -point strips. Line fuse is
mounted on rear apron of chassis.

The Power Supply

The high current power
supply uses 300 µµfd.

filter capacitors and 5 ohm protective resistors. R -C decoupling is used to minimize hum
in the low level audio stages. As with all
"power- transformerless" equipment, care must
be taken when connecting this amplifier to
other pieces of equipment to ensure that the
grounded side of the power line is connected
to the chassis. This may be achieved by the
use of a polarized line plug, or a small isolation transformer may be employed.
The Equalizing

Circuit

As would be expected, 40

db of feedback can only be
applied within a loop having a minimum of phase shift or circuit instability will result. Since the loudspeaker

O

www.americanradiohistory.com

HANDBOOK

Transformerless Amplifier

o

149

.0

0.8

0 6

-10

0.
0.2
20

10

100

1000

10 KC

FREQUENCY

100 KC

1000 KC

Figure 29.

The balance adjustment for zero d.c. current through the speaker voice coil can be
made with a milliammeter in series with the
coil, or by measuring the voltage across the
coil with a sensitive voltmeter.
Amplifier

The amplifier is built upon
an aluminum chassis measuring 8" x 10" x 2 ". Perforated end pieces and 1/4 -inch holes drilled
around the 6082 tube sockets insure adequate
ventilation. Layout of the major components
is shown in figure 26, and placement of the
under -chassis components is shown in figure
28. As no a.c. power transformer is used,
ground currents are of small concern, and the
ground bus wiring technique need not be emConstruction

20

25

POWER OUTPUT (wnrrs)

A- Overall frequency response of amplifier
B-Distortion versus power output of amplifier

impedance becomes inductive above the audio
range it causes an increase in phase shift and
loop gain. To avoid instability an impedance
can be shunted across the voice coil to prevent
the output reactance from rising at the higher
audio frequencies. Three networks that have
been used successfully for this purpose are
shown in figure 27. The 180 ohm res for
merely limits the maximum impedance of the
output system and thus preven , excessive
feedback. The 0.5 pfd. capacitor places a low
impedance across the inductive load which is
effective at the higher audio frequencies. The
series 16 ohm resistor and 0.01 pfd. capacitor
places a resistance across the speaker at the
higher frequencies and an open circuit at the
lower frequencies. This serves to provide constant impedance and feedback over the frequency range of the amplifier.

15

10

o

0

ployed. In its place, a tinned copper wire is
run between the various chassis ground points.
Ground connections may now be made to the
socket grounding lugs, or to terminal strip
ground points. A.c. filament and power leads
are twisted wherever possible, and are run
around the outer edges of the chassis.
Point -to -point wiring technique is used,
with small capacitors and resistors mounted
to socket pins or to phenolic tie -point strips
placed near the sockets. The small silicon rectifiers are mounted to tie -point strips placed
near the upright filter capacitors.
Several of the filter capacitors do not have
their negative terminal at ground potential.
It is therefore necessary to mount the capacitor
on a phenolic plate and to slip a fiber insulating jacket over the metal shell.

Amplifier

The frequency response of
the amplifier is flat within
one db from 10 cycles to over
100 kilocycles. Since R -C coupled circuits are
used throughout, there is no serious limitation
on frequency response, and the response is
down only 4 db at 250,000 cycles. The inter stage coupling networks limit the low frequency response below 10 cycles.
Harmonic distortion and intermodulation at
full rated output are exceptionally low and virtually independent of frequency. The ability
to deliver 25 watts at 20 cycles and below
with negligible distortion is practically impossible in a transformer -type circuit of similar
mid -frequency power rating. Square wave response of the amplifier as measured between
20 cycles and 50 kilocycles is extremely good.
Performance

www.americanradiohistory.com

--IilIFte
LO-russ

T seCT,oM

LOIS

r4ss i/

011411

..,f

7T

5e0110M

VALUES

SCALE
FREQUENCY
HIGH -PASS
LOW -PASS
.00

j-----;

M1oM-491 T seCT,ON

seCT,OM

L

LOAD RESISTANCE

C

25.0

90

1000

10000

1100

9000

200
80

20.0

70

5

.5

1300

6

6

r

.7

1400
1500

.6

e

60

15.0

9

.9

10

1.0

1600
1700

W

tu

50

1600
1900

e000

Z-

--r

1000

6000
W
-1

u

5000 n

2000

M

4.s

W
40

10.0

V
Z 2500
<

2.0

O

4000

r
u

9.0

O 3000
Z

3.0

e.0

3000

LC

7.0

40

.0

Ñ

50

5.0

u 4000

60

e.0

1h.
W

2

6.0
70

90

5.0

20

7.0

I

e0

100

9.0

10.0

4.0

--i-

-

J

= 5000_4r
3
ta 1000 -- EA
Ñ

8.0

I

-5.-

15.0

C 1000

00

20.0

8000

300

30.0

150

3.0

J9000

2.5

10

10000

For both

Pi -type

1000

-

--

Courler, rrcdic e.0,9 9ni,
FILTER DESIGN

2000

n

1000

ing

CO.

CHART

and T -type Sections

connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right side scale for high -pass) with load on left -hand side of right -hand scale by means of a straight -edge.
Then read the value of L from the point where the edge intersects the left side of the center scale. Readings are in henries for frequencies in cycles per second.
To

find

L,

To find C, connect cut -off frequency on left -hand scale (using left -side scale for low -pass and right side scale for high pass) with the load on the right -hand side of the right -hand scale. Then read the
value of C from the point where the straightedge cuts the right side of the center scale. Readings are

in microfarads for frequencies in cycles per

d.

For frequencies in kilocycles, C is expressed in thousands of micromicrofarads, L is expressed in
mlllihenries. For frequencies in megacycles, L is expressed in microhenries and C is expressed in micromlcrofarads.
For each tenfold increase In the value of load resistance multiply L by 10 and divide C by
For each ten fold decrease in frequency multiply L by 10 end multiply C by 10.

150
www.americanradiohistory.com

IO.

CHAPTER EIGHT

Radio Frequency
Vacuum Tube Amplifiers

TUNED RF VACUUM TUBE AMPLIFIERS
Tuned r -f voltage amplifiers are used in receivers for the amplification of the incoming
r -f signal and for the amplification of intermediate frequency signals after the incoming
frequency has been converted to the intermediate frequency by the mixer stage. Signal frequency stages are normally called tuned r -f
amplifiers and intermediate -frequency stages
are called i-f amplifiers. Both tuned r -f and
i -f amplifiers are operated Class A and normally operate at signal levels from a fraction
of a microvolt to amplitudes as high as 10 to
50 volts at the plate of the last i -f stage in a
receiver.

first tuned circuit due to its equivalent coupled resistance at resonance. The noise voltage generated due to antenna radiation resistance and to equivalent tuned circuit resistance
is similar to that generated in a resistor due
to thermal agitation and is expressed by the
following equation:
En'

k

=

R =

Grid Circuit

Considerations

1f =
Since the full amplification of a receiver follows the first tuned circuit, the operating conditions existing in that circuit and in its coupling to the antenna on one side and to the
grid of the first amplifier stage on the other
are of greatest importance in determining the
signal -to -noise ratio of the receiver on weak

signals.
highest
ratio of signal -to -noise be impressed on the grid of the first
r -f amplifier tube. Attaining the optimum ratio
is a complex problem since noise will be generated in the antenna due to its equivalent
radiation resistance (this noise is in addition
to any noise of atmospheric origin) and in the
First Tuned
Circuit

It is obvious that the

4kTRAf

Where: E° = r -m -s value of noise voltage over
the interval .1f

T =
8 -1

=

Boltzman's constant = 1374
X 10-22 joule per °K.
Absolute temperature °K.
Resistive component of impedance across which thermal noise
is developed.
Frequency band across which
voltage is measured.

In the above equation \f is essentially the
frequency band passed by the intermediate frequency amplifier of the receiver under consideration. This equation can be greatly simplified for the conditions normally encountered
in communications work. If we assume the following conditions: T = 300° K or 27° C or
80.5° F, room temperature; 1f = 8000 cycles
(the average pass band of a communications
receiver or speech amplifier) the equation remicrovolts. Acduces to: Et.m.s. = 0.0115
cordingly, the thermal -agitation voltage appearing in the center of half -wave antenna (assuming effective temperature to be 300° K)
having a radiation resistance of 73 ohms is

151

www.americanradiohistory.com

152

R

-F

Vacuum

Tube

Amplifiers

approximately 0.096 microvolts. Also, the thermal agitation voltage appearing across a 500,000 -ohm grid resistor in the first stage of a
speech amplifier is approximately 8 microvolts
under the conditions cited above. Further, the
voltage due to thermal agitation being impressed on the grid of the first r -f stage in a
receiver by a first tuned circuit whose resonant resistance is 50,000 ohms is approximately
2.5 microvolts. Suffice to say, however, that
the value of thermal agitation voltage appearing across the first tuned circuit when the antenna is properly coupled to this circuit will
be very much less than this value.
It is common practice to match the impedance of the antenna transmission line to the
input impedance of the grid of the first r -f amplifier stage in a receiver. This is the condition of antenna coupling which gives maximum
gain in the receiver. However, when u -h -f tubes
such as acorns and miniatures are used at frequencies somewhat less than their maximum
capabilities, a significant improvement in signal -to -noise ratio can be attained by increasing the coupling between the antenna and first
tuned circuit to a value greater than that which
gives greatest signal amplitude out of the receiver. In other words, in the 10, 6, and 2 meter bands it is possible to attain somewhat improved signal -to -noise ratio by increasing antenna coupling to the point where the gain of
the receiver is slightly reduced.
It is always possible, in addition, to obtain
improved signal -to -noise ratio in a v -h -f receiver through the use of tubes which have
improved input impedance characteristics at
the frequency in question over conventional
types.
The limiting condition for sensitivity in any receiver is the
thermal noise generated in the antenna and in
the first tuned circuit. However, with proper
coupling between the antenna and the grid of
the tube, through the first tuned circuit, the

Noise Factor

noise contribution of the first tuned circuit
can be made quite small. Unfortunately, though,
the major noise contribution in a properly designed receiver is that of the first tube. The
noise contribution due to electron flow and
due to losses in the tube can be lumped into
an equivalent value of resistance which, if
placed in the grid circuit of a perfect tube having the same gain but no noise would give the
same noise voltage output in the plate load.
The equivalent noise resistance of tubes such
as the 6SK7, 6SG7, etc., runs from 5000 to
10,000 ohms. Very high Gm tubes such as the
6AC7 and 6AK5 have equivalent noise resistances as low as 700 to 1500 ohms. The lower
the value of equivalent noise resistance, the

THE

RADIO

lower will be the noise output under a fixed
set of conditions.
The equivalent noise resistance of a tube
must not be confused with the actual input
loading resistance of a tube. For highest signal -to -noise ratio in an amplifier the input
loading resistance should be as high as possible so that the amount of voltage that can be
developed from grid to ground by the antenna
energy will be as high as possible. The equivalent noise resistance should be as low as
possible so that the noise generated by this
resistance will be lower than that attributable
to the antenna and first tuned circuit, and the
losses in the first tuned circuit should be as
low as possible.
The absolute sensitivity of receivers has
been designated in recent years in government
and commercial work by an arbitrary dimensionless number known as "noise factor" or N.
The noise factor is the ratio of noise output
of a "perfect" receiver having a given amount
of gain with a dummy antenna matched to its
input, to the noise output of the receiver under
measurement having the same amount of gain
with the dummy antenna matched to its input.
Although a perfect receiver is not a physically
realizable thing, the noise factor of a receiver
under measurement can be determined by calculation from the amount of additional noise
(from a temperature -limited diode or other calibrated noise generator) required to increase
the noise power output of a receiver by a predetermined amount.
Tube Input

As has been mentioned in a pre -

vious paragraph, greatest gain
in a receiver is obtained when
the antenna is matched, through the r -f coupling transformer, to the input resistance of
the r -f tube. However, the higher the ratio of
tube input resistance to equivalent noise resistance of the tube the higher will be the signal -to -noise ratio of the stage -and of course,
the better will be the noise factor of the overall receiver. The input resistance of a tube
is very high at frequencies in the broadcast
band and gradually decreases as the frequency
increases. Tube input resistance on conventional tube types begins to become an important factor at frequencies of about 25 Mc. and
above. At frequencies above about 100 Mc. the
use of conventional tube types becomes impracticable since the input resistance of the
tube has become so much lower than the equivalent noise resistance that it is impossible
to attain reasonable signal -to -noise ratio on
any but very strong signals. Hence, special
v -h-f tube types such as the 6AK5, 6ÁG5, and
6CB6 must be used.
The lowering of the effective input resistLoading

www.americanradiohistory.com

HANDBOOK

R

ance of a vacuum tube at higher frequencies
is brought about by a number of factors. The
first, and most obvious, is the fact that the
dielectric loss in the internal insulators, and
in the base and press of the tube increases
with frequency. The second factor is due to
the fact that a finite time is required for an
electron to move from the space charge in the
vicinity of the cathode, pass between the grid
wires, and travel on to the plate. The fact that
the electrostatic effect of the grid on the moving electron acts over an appreciable portion
of a cycle at these high frequencies causes a
current flow in the grid circuit which appears
to the input circuit feeding the grid as a resistance. The decrease in input resistance of
a tube due to electron transit time varies as
the square of the frequency. The undesirable
effects of transit time can be reduced in certain cases by the use of higher plate voltages.
Transit time varies inversely as the square
root of the applied plate voltage.
Cathode lead inductance is an additional
cause of reduced input resistance at high frequencies. This effect has been reduced in certain tubes such as the 6S117 and the 6AK5 by
providing two cathode leads on the tube base.
One cathode lead should be connected to the
input circuit of the tube and the other lead
should be connected to the by -pass capacitor
for the plate return of the tube.
The reader is referred to the Radiation Laboratory Series, Volume 23: "Microwave Receivers" (McGraw -Hill, publishers) for additional
information on noise factor and input loading
of vacuum tubes.

Amplifiers

-F

153

OA

AMPLIFICATION AT RESONANCE (APPROX.) =GMWLQ

OB

AMPLIFICATION AT RESONANCE (APPROX ) =GWMQ

© AMPLIFICATION

AT RESONANCE(APPRO[kGMK

U)

K2t-1-s
1

QP

WHERE

S

PRI. ANO SEC. RESONANT AT SAME FREQUENCY
2 K IS COEFFICIENT OF COUPLING
1.

IF FRI. AND SEC. Q ARE APPROXIMATELY THE SAME.

8 -2

TOTAL BANDWIDTH
1.2 K
CENTER FREQUENCY
MAXIMUM AMPLITUDE OCCURS AT CRITICAL COUPLING

Plate- Circuit
Considerations

WHEN

-

KQP

Noise is generated in a vacuum tube by the
fact that the current flow within the tube is not
a smooth flow but rather is made up of the continuous arrival of particles (electrons) at a
very high rate. This shot effect is a source of
noise in the tube, but its effect is referred
back to the grid circuit of the tube since it is
included in the equivalent noise resistance
discussed in the preceding paragraphs.
For the purpose of this section,
it will be considered that the
function of the plate load circuit of a tuned vacuum -tube amplifier is to deliver energy to the next stage with the greatest
Plate Circuit
Coupling

efficiency over the required band of frequencies. Figure 1 shows three methods of inter stage coupling for tuned r -f voltage amplifiers.
In figure IA omega (w) is 2n times the resonant frequency of the circuit in the plate of

Figure
Gain

1

equations for pentode r -f amplifier
stages operating into a tuned load

the amplifier tube, and L and Q are the inductance and Q of the inductor L. In figure 1B the
notation is the same and M is the mutual inductance between the primary coil and the secondary coil. In figure 1C the notation is again
the same and k is the coefficient of coupling
between the two tuned circuits. As the coefficient of coupling between the circuits is
increased the bandwidth becomes greater but
the response over the band becomes progressively more double -humped. The response over
the band is the most flat when the Q's of primary and secondary are approximately the same
and the value of each Q is equal to 1.75/k.

www.americanradiohistory.com

154

R

-F

Variable -Mu Tubes

Vacuum

Tube

Amplifiers

It is common practice to

control the gain of a succession of r -f or i -f amplifier stages by varying the average bias on
their control grids. H)wever, as the bias is
raised above the operating value on a conventional sharp- cutoff tube the tube becomes increasingly non -linear in operation as cutoff of
plate current is approached. The effect of such
non -linearity is to cause cross modulation between strong signals which appear on the grid
of the tube. When a tube operating in such a
manner is in one of the first stages of a receiver a number of signals are appearing on its
grid simultaneously and cross modulation between them will take place. The result of this
effect is to produce a large number of spurious
signals in the output of the receiver -in most
in

R

-F Stages

THE

RADIO

cases these signals will carry the modulation
of both the carriers which have been cross
modulated to produce the spurious signal.
The undesirable effect of cross modulation
can be eliminated in most cases and greatly
reduced in the balance through the use of a
variable -mu tube in all stages which have a -v-c
voltage or other large negative bias applied to
their grids. The variable -mu tube has a characteristic which causes the cutoff of plate current to be gradual with an increase in grid
bias, and the reduction in plate current is accompanied by a decrease in the effective amplification factor of the tube. Variable -mu tubes
ordinarily have somewhat reduced Gm as compared to a sharp- cutoff tube of the same group.
Hence the sharp- cutoff tube will perform best
in stages to which a-v -c voltage is not applied.

RADIO- FREQUENCY POWER AMPLIFIERS
All modern transmitters in the medium -frequency range and an increasing percentage of
those in the v -h -f and u -h-f ranges consist of
a comparatively low -level source of radio-frequency energy which is multiplied in frequency
and successively amplified to the desired power
level. Microwave transmitters are still predominately of the self- excited oscillator type, but
when it is possible to use r -f amplifiers in
s -h -f transmitters the flexibility of their application will be increased. The following portion of this chapter will be devoted, however,
to the method of operation and calculation of
operating characteristics of r-f power amplifiers for operation in the range of approximately 3.5 to 500 Mc.
8 -3

Class C R -F
Power Amplifiers

The majority of r -f power amplifiers fall into
the Class C category since such stages can
be made to give the best plate circuit efficiency of any present type of vacuum-tube amplifier. Hence, the cost of tubes for such a stage
and the cost of the power to supply that stage
is least for any given power output. Nevertheless, the Class C amplifier gives less power
gain than either a Class A or Class B amplifier under similar conditions since the grid of
a Class C stage must be driven highly positive over the portion of the cycle of the exciting wave when the plate voltage on the amplifier is low, and must be at a large negative
potential over a large portion of the cycle so

that no plate current will flow except when
plate voltage is very low. This, in fact, is the
fundamental reason why the plate circuit efficiency of a Class C amplifier stage can be
made high -plate current is cut off at all times
except when the plate -to- cathode voltage drop
across the tube is at its lowest value. Class
C amplifiers almost invariably operate into a
tuned tank circuit as a load, and as a result
are used as amplifiers of a single frequency
or of a comparatively narrow band of frequencies.
2 shows the relation ships between the various
voltages and currents over
one cycle of the exciting grid voltage for a
Class C amplifier stage. The notation given in
figure 2 and in the discussion to follow is the
same as given at the first of Chapter Six under "Symbols for Vacuum -Tube Parameters."
The various manufacturers of vacuum tubes
publish booklets listing in adequate detail alternative Class C operating conditions for the
tubes which they manufacture. In addition,
operating condition sheets for any particular
type of vacuum tube are available for the asking from the different vacuum -tube manufacturers. It is, nevertheless, often desirable to
determine optimum operating conditions for a
tube under a particular set of circumstances.
To assist in such calculations the following
paragraphs are devoted to a method of calculating Class C operating conditions which is
moderately simple and yet sufficiently accurate for all practical purposes.

Relationships in
Class

C

Stage

www.americanradiohistory.com

Figure

Class

HANDBOOK

R

C

-F Amplifiers

155

tional grid voltage -plate current operating
curves, the calculation is considerably simplified if the alternative "constant- current

curve" of the tube in question is used. This is
true since the operating line of a Class C amplifier is a straight line on a set of constantcurrent curves. A set of constant -current curves
on the 250TH tube with a sample load line
drawn thereon is shown in figure 5.
In calculating and predicting the operation

PLATE
VOLTAGE
EPM

EBB

P

eP

O1._.L.L- --'-- -1-I

--- r

I

1

PEAK

PLATE
CURRENT

_

I

If--t- - -

--I--

-I

I19P-.1.-9P+11

I

I

I

I

I

I

7I

I

I

t
I

11
VI

1

I

r
I

I
EGM

e

-

I

I

II
II

I

I

I

I

-I

I

-

V

t1
III

-

I

III
11

III

0i- d11i
Ecc

-i

FUNDAMENTAL COMPONENT
OF PLATE CURRENT

'I

I

IG MAS.

1

:
I

{

-1-,,,*

I

I

-

I

I

I

1------ I-

I11

I

III
111

I
I

III
GRID
III VOLTAGE
11-- -I
G

'-Ieca-1--

Figure
Instantaneous electrode
voltages and currents for

I

2

and
a

amplifier

Calculation of Class
C Amplifier Operating
Characteristics

GRID

I

FI-CURRENT

tank

Class

circuit

C r -f

power

Although Class C opcrating conditions can
be determined with the
aid of the more conven-

of a vacuum tube as a Class C radio -frequency
amplifier, the considerations which determine
the operating conditions are plate efficiency,
power output required, maximum allowable
plate and grid dissipation, maximum allowable
plate voltage and maximum allowable plate
current. The values chosen for these factors
will depend both upon the demands of a par-

ticular application and upon the tube chosen.
The plate and grid currents of a Class C
amplifier tube are periodic pulses, the durations of which are always less than 180 degrees. For this reason the average grid current, average plate current, power output, driving power, etc., cannot be directly calculated
but must be determined by a Fourier analysis
from points selected at proper intervals along
the line of operation as plotted upon the constant- current characteristics. This may be done
either analytically or graphically. While the
Fourier analysis has the advantage of accuracy, it also has the disadvantage of being
tedious and involved.
The approximate analysis which follows
has proved to be sufficiently accurate for most
applications. This type of analysis also has
the advantage of giving the desired information at the first trial. The system is direct in
giving the desired information since the important factors, power output, plate efficiency,
and plate voltage are arbitrarily selected at
the beginning.

first step in the method to
described is to determine the
power which must be delivered
by the Class C amplifier. In making this determination it is well to remember that ordinarily
from 5 to 10 per cent of the power delivered
by the amplifier tube or tubes will be lost in
well- designed tank and coupling circuits at
frequencies below 20 Mc. Above 20 Mc. the
tank and circuit losses are ordinarily somewhat above 10 per cent.
The plate power input necessary to produce
the desired output is determined by the plate
efficiency: Pin = Pout/Np.
For most applications tt is desirable to operate at the highest practicable efficiency. High efficiency operation usually requires less expensive tubes and power supplies, and the
Method of

The

Calculation

be

www.americanradiohistory.com

156

R

-F

Vacuum

Tube

\

Amplifiers

THE

i'\ E
\\
agi
E \EE
EE

\u
=\

MENE

7.0

E

.0

E

\1

O

7.0

E

EEI \

.0

40

30

RADIO

tt

e

RATIO

,.

iáu

t

3.0

I

-10

EEE IMEN
-20

-10

.1

RATIO

Figure 3
Relationship between the peak value of the
fundamental component of the tube plate current, and average plate current; as compared
to the ratio of the instantaneous peak value
of tube plate current, and average plate
current

amount of artificial cooling required is frequently less than for low- efficiency operation.
On the other hand, high- efficiency operation
usually requires more driving power and involves the use of higher plate voltages and
higher peak tube voltages. The better types
of triodes will ordinarily operate at a plate
efficiency of 75 to 85 per cent at the highest
rated plate voltage, and at a plate efficiency
of 65 to 75 per cent at intermediate values of

plate voltage.
The first determining factor in selecting a
tube or tubes for a particular application is
the amount of plate dissipation which will be
required of the stage. The total plate dissipation rating for the tube or tubes to be used in
the stage must be equal to or greater than that
calculated from: Pp = Pin - Pout.
After selecting a tube or tubes to meet the
power output and plate dissipation requirements it becomes necessary to determine from
the tube characteristics whether the tube selected is capable of the desired operation and,
if so, to determine the driving power, grid
bias, and grid dissipation.
The complete procedure necessary to determine a set of Class C amplifier operating conditions is given in the following steps:
1. Select the plate voltage, power output,
and efficiency.

Figure

4

Relationship between the ratio of the peak
value of the fundamental component of the
grid excitation voltage, and the overage grid
bias; as compared to the ratio between instantaneous peak grid current and average
grid current

2.

Determine plate input from: Pin

=

Pout/Np.

3 Determine plate dissipation
Pp= Pin - Pout Pp must

from:

not exceed

maximum rated plate dissipation for tube
or tubes selected.

4. Determine average plate current from:
lb = Pin /Ebb
5.

Determine approximate

;p.a.

from:

4.9 lb for Np = 0.85
tpmas 4.5 lb for Np = 0.80
tpmas = 4.0 'b for N = 0.75
tpmax= 3.51b for Np =0.70
tpmax

6. Locate

=
=

the point on

constant -current

characteristics where the constant plate

current line corresponding to the approximate ipmax determined in step 5
crosses the line of equal plate and grid
voltages (diode line). Read epmin at this
point. In a few cases the lines of constant plate current will inflect sharply
upward before reaching the diode line.
In these cases epmin should not be read
at the diode line but at the point where
the plate current line intersects a line
drawn from the origin through these
points of inflection.

www.americanradiohistory.com

FIRST TRIAL POINT

-s,

N
Om

157

Constant Current Calculations

HANDBOOK

FINAL POINT

EIMAC 250TH
CONSTANT CURRENT
CHARACTERISTICS
....

sa:

pP:

.
,

r-;pze

o

ó

-_-.

00

_

Ti

ERE

7b.
....... .........

,
MOO

EGO=

- 240

LOAD LINE

x00

XOD

Ebb =4-3500

PLATE VOLTAGE -VOLTS

FIGURE

5

Active portion of the operating load line for an Eimoc 250TH Class C r -f power amplifier,
showing first trial point and the final operating point

7. Calculate Epm from: Epm = Ebb

- epmin

13.

8. Calculate the ratio Ipm /lb from:
1pm

lb

2

Ecc

1

Epm

Fos

Calculate a new value for ipmax from
the ratio found in step 9.
tpm as = (ratio from step 9) lb

Ecc

f3p

=2.32(

Ipm
Ib

-

-

µ

egmp)

Ebb

fi

1

X

L

for tetrodes, where

camp cos

O

-

En,

J
1112

tt

is the grid- screen
amplification factor, and Ec2 is the d -c
screen voltage.

and 10.

cos

\

1- cos 6p

constant current characteristics for the values of
epmin and ipmax determined in steps 6

Calculate the cosine of one -half the
angle of plate current flow from:

Epm
Op

X

- cos Op

for triodes.

11. Read egmp and igmax from the

12.

1

-

Np Ebb

9. From the ratio of Ipm /Ib calculated in
step 8 determine the ratio ipmax /Ib from
figure 3.
10.

Calculate the grid bias voltage from:

14.

Calculate the peak fundamental grid excitation voltage from:
Earn = egmp

-

Ecc

1.57)
15.

Calculate the ratio Egm /Ecc for the val-

www.americanradiohistory.com

158

R

-F

Vacuum

Amplifiers

Tube

ues of Ecc and Egm found in steps 13

12.

and 14.

Bp=2.32(1.73-1.57)=0.37

ratio
13.

1

-

- 0.37

X

(_ 3240

[0.37

3500

240)

37

37

14.

- 240 volts
Egm = 240 - ( -240) = 480 volts grid

15.

Egm /Ecc =

16.

igmax /Ic = 5.75 (from figure 4)

17.

le = 0.430/5.75 = 0.075 amp. (75 ma.
grid current)

18.

Pd = 0.9X480X0.075 = 32.5 watts
driving power

=

tgmax
Ratio from step

Ecc

1

Calculate the average grid current from
the ratio found in step 16, and the value
of igmax found in step 11:
IC

cos

RADIO

(9p = 68.3°)

16. Read igmax /Ic from figure 4 for the
Egm /Ecc found in step 15.

17.

THE

16

swing
18. Calculate approximate grid driving pow-

er from:

Pd

0.9 Egmlc

=

Calculate grid dissipation from:
Pa = Pa + Eccic

19.

Pg must not exceed the maximum rated

grid dissipation for the tube selected.

Sample

typical example of a Class C
amplifier calculation is shown
A

in the example below. Reference
3, 4 and 5 in the calcula-

is made to figures
tion.
1.

Desired power output -800 watts.

2.

Desired plate voltage -3500 volts.
Desired plate efficiency -80 per cent
(Np = 0.80)
Pin = 800 /0.80 = 1000 watts

3.

The power output of any type of r -f ampli-

fier is equal to:

Ib

-

It is frequently of importance to know the
value of load impedance into which a Class
C amplifier operating under a certain set of
conditions should operate. This is simply R L_
Epm /Ipm. In the case of the operating conditions just determined for a 250TH amplifier
stage the value of load impedance is:
RL _

Approximate ipmax = 0.285 X 4.5
= 1.28 ampere

trial point)
7.

Epm = 3500

volts (see figure

- 260 = 3240

5

11.

egmp = 240 volts
igmax = 0.430 amperes

3240

Ipm

.495

6600 ohms

-Xlb
lb

of Amplifier
Tank Circuit

volts

(Both above from final point on figure

Epm

Ipm =

In order to obtain good plate
tank circuit tuning and low
radiation of harmonics from
an amplifier it is necessary that the plate tank
circuit have the correct Q. Charts giving compromise values of Q for Class C amplifiers
are given in the chapter, Generation o/ R -F
Energy. However, the amount of inductance

Q

9. ipmax /Ib = 4.1 (from figure 3)

ipmax = 0.285X4.1 = 1.17

-- -Ipm

first

8. Ipm /lb = 2X0.80X3500/3240 =
5600/3240 = 1.73

10.

-

ratio determined in step 8 above (in this type
of calculation) by multiplying this ratio times

Pp = 1000
800 = 200 watts
Use 250TH; max. Pp = 250w;µ = 37.

6. epmin = 260

2

IpmEpm /2 = Po
Ipm can be determined, of course, from the

4. lb = 1000/3500 = 0.285 ampere (285 ma.)
Max. Ib for 250TH is 350 ma.
5.

- 240 = -

(- 240X0.75) = 14.5 watts
grid dissipation
Max. Pg for 250TH is 40 watts

19. Pg = 32.5

Calculation

480/

5)

required for a specified tank circuit Q under
specified operating conditions can be calculated from the following expression:

www.americanradiohistory.com

Class

HANDBOOK

R

B

Q

Quick Method of
Calculating Amplifier
Plate Efficiency

The plate circuit efficiency of a Class B or
Class C r -f amplifier
can be determined from

the following facts. The plate circuit efficiency
of such an amplifier is equal to the product of
two factors,
which is equal to the ratio of
Epm to Ebb (F, = Epm /Ebb) and
which is
proportional to the one -half angle of plate current flow, 0p. A graph of F, against both 0,
and cos Bp is given in figure 6. Either 0p or
cos Bp may be used to determine F,. Cos 0p
may be determined either from the procedure
previously given for making Class C amplifier
computations or it may be determined from the
following expression:

F

F

Bp =

-

Ecc

+

Ebb

µEBm-Epm
Example of
Method

is desired to know the one -half
angle of plate current flow and
It

the plate circuit efficiency for
an 812 tube operating under the following conditions which have been assumed from inspection of the data and curves given in the RCA
Transmitting Tube Handbook HB -3:
1.

Ebb = 1100 volts
Ecc = -40 volts

1.'

....,....
.......,....
.......-....
...\\...
Ism

= 2 ir X operating frequency
= Tank inductance
= Required tube load impedance
= Effective tank circuit Q

tank circuit Q of 12 to 20 is recommended
for all normal conditions. However, if a balanced push -pull amplifier is employed the tank
receives two impulses per cycle and the circuit Q may be lowered somewhat from the
above values.

cos

DN,IIII.`....MM....

MN

A

159

R\......

Ri,

L
RL
Q

Amplifiers

-F

aee1
F2

aee
o.

i.........,,..
I..........,.

awl
D

ore
ore

ar211..........,
.... .....
aM

0

,

10

20

30

PP

IN

40

50

e0

70

e0

90

100 110

120

ELECTRICAL DEGREES
. .». <.. .... .
.
.:,. ... <.,. >... -..,»
I

cos

I

AP

Figure 6
Relationship between Factor F_ and the
half -angle of plate current flow in an amplifier with sine -wave input and output voltage,
operating at a grid -bias voltage greater than
cut -off

5. Np= F, X F, = 0.91 X 0.79 = 0.72

(72 per cent efficiency)
F, could be called the plate -voltage-swing
efficiency factor, and F2 can be called the
operating -angle efficiency factor or the maximum possible efficiency of any stage running
with that value of half -angle of plate current
flow.
Np is, of course, only the ratio between
power output and power input. If it is desired
to determine the power input, exciting power,
and grid current of the stage, these can be obtained through the use of steps 7, 8, 9, and 10
of the previously given method for power inis
put and output; and knowing that
0.095 ampere the grid circuit conditions can
be determined through the use of steps 15, 16,
17, 18 and 19.

= 29

Ea.

= 120 volts
Epm = 1000 volts

2.

F,

3.

cos 0p=

=

Epm/Ebb

=

8 -4

0.91

- 29 X 40 + 1100
29 X 120 - 1000

60

- 0.025

2480
4.

F2= 0.79 (by reference to figure 6)

Class B Radio
Frequency Power Amplifiers

Radio frequency power amplifiers operating
under Class B conditions of grid bias and excitation voltage are used in two general types
of applications in transmitters. The first general application is as a buffer amplifier stage
where it is desired to obtain a high value of
power amplification in a particular stage. A
particular tube type operated with a given
plate voltage will be capable of somewhat
greater output for a certain amount of excitation power when operated as a Class B ampli-

www.americanradiohistory.com

160

R

-F

Tube

Vacuum

Amplifiers
Ecz= 7-400

l

RADIO

THE
V.

Ec3=0v.
Eci=

II

Ec=+Go

__A

___

rd.

1

}s0

Eci=+4o

i

I

\t

Eci= +20

Eci=

I

1

i

I

I

iii

/

c2,Ec=+ioo

J--

Icz Ec=+w

l00

200

300

400

I

Fri,

_,--.-500

G00

700

-20

Eci=-4o
G0o

am

000

um,

PLATE VOLTS

.Hr,

v.

A.....

._....

..,.,.

...

Figure 7
AVERAGE PLATE CHARACTERISTICS OF 813 TUBE

fier than when operated as
fier.

a

Class C ampli-

Calculation of

Calculation of the operating
conditions for this type of
Characteristics
Class B r -f amplifier can be
carried out in a manner similar to that described in the previous paragraphs, except that the grid bias voltage is set
on the tube before calculation at the value:
Ecc =
Ebb /IL. Since the grid bias is set at
cutoff the one-half angle of plate current flow
is 900; hence cos
is fixed at 0.00. The
plate circuit efficiency for a Class B r -f amplifier operated in this manner can be determined in the following manner:
Operating

NP

=78.5

(

-1
Epm

"Class
Linear"

B

plication.
Calculation of Operoting Parameters for a
Class B Linear Amplifier

Ebb //

The

calculated. Then, with the exciting voltage
reduced to one -half for the no- modulation condition of the exciting wave, and with the same
value of load resistance reflected on the tube,
the plate input and plate efficiency will drop
to approximately one-half the values at the
100 per cent positive modulation peak and the
power output of the stage will drop to onefourth the peak- modulation value. On the negative modulation peak the input, efficiency, and
output all drop to zero.
In general, the proper plate voltage, bias
voltage, load resistance and power output
listed in the tube tables for Class B audio
work will also apply to Class B linear r -f ap-

The second type of Class B
r-f amplifier is the so- called

Class Il linear amplifier which
is often used in transmitters for the amplification of a single - sideband signal or a conventional amplitude- modulated wave. Calculation
of operating conditions may be carried out in
a manner similar to that previously described
with the following exceptions: The first trial
operating point is chosen on the basis of the
100 per cent positive modulation peak of the
modulated exciting wave. The plate circuit
and grid peak voltages and currents can then
be determined and the power input and output

7 illustrates
characteristic
curves for an 813

Figure

the

tube.

Assume

the

plate supply to be 2000 volts, and the screen
supply to be 400 volts. To determine the operating parameters of this tube as a Class B linear r -f amplifier, the following steps should
be taken:
1.

The grid bias is chosen so that the resting plate current will produce approximately 1/3 of the maximum plate dissipation of the tube. The maximum dissipation of the 813 is 125 watts, so the
bias is set to allow one -third of this
value, or 42 watts of resting dissipation.
At a plate potential of 2000 volts, a

www.americanradiohistory.com

HANDBOOK

2.

Linear Amplifier Parameters

plate current of 21 milliamperes will
produce this figure. Referring to figure
7, a grid bias of -45 volts is approximately correct.
A practical Class 13 linear r -f amplifier
runs at an efficiency of about 66% at full
output, the efficiency dropping to about
33% with an unmodulated exciting signal. In the case of single- sideband suppressed carrier excitation, a no- excitation condition is substituted for the unmodulated excitation case, and the linear amplifier runs at the resting or quiescent input of 42 watts with no exciting
signal. The peak allowable power input
to the 813 is:
Input Peak Power (Wp) _
(watts)
Plate Dissipation X 100
(100

-

BO

`

GO

Ec3=ov.

Eci=+ioov.

ci=rsov.,

40

¡ Ec-raov.
Rib_ Eu=+sov

xo

Ecr+20
o

-

_

-

.

200

100

Eg

VS.

E

x

P

Ep

-

400

_

x

-= 0.189 ampere

2000

0.189

1580

=

=

0.5 x .189
6000 ohms

9. If a loaded plate tank circuit Q of 12 is

desired, the reactance of the plate tank
capacitor at the r e s on an t frequency

The plate current flow of the linear amplifier is 1800, and the plate current
pulses have a peak of 3.14 times the
maximum signal current:
3.14

resistance is:
epmin

379

Wp

Ep

-

0.5ipmax

The maximum signal plate current is:
tpmax =

should be:

Reactance (ohms) =

0.595 ampere

RL
--

Referring to figure 7, a current of 0.605
ampere (Point A) will flow at a positive
grid potential of 60 volts and a minimum
plate potential of 420 volts. The grid is
biased at -45 volts, so a peak r -f grid
voltage of 60+45 volts = 105 volts is required.
The grid driving power required for the
B linear stage may be found by the
aid of figure 8. It is one -quarter the product of the peak grid current times the
peak grid voltage:

Class

0.02

X 105

Pp =

10.

500 ohms

For an operating frequency of 4.0 Mc.,

the effective resonant capacity is:
106
C=

=

6.28
11.

x

4.0

x

80 µµtd.

500

The inductance required to resonate at
4.0 Mc. with this value of capacity is:
500
L

- 0.53

6000
=

12

Q

6.

300

Figure 8
CHARACTERISTICS OF 813
TUBE

8. The plate load

100 = 379 watts

RL

5.

E

PLATE VOLTS

33

4.

ECO +400 V.

-% plate efficiency)

125

3.

161

watt

6.28

=

x

4.0

19.9 microhenries

4

7.

The single tone power output of the 813
stage is:
Pp = 78.5 (Ep - epmin) x Ip
Pp = 78.5 (2000 - 420) x .189 = 235 watts

Grid Circuit

The maximum positive grid
potential is 60 volts, and
the peak r -f grid voltage is
105 volts. Required driving power is 0.53 watt.
The equivalent grid resistance of this stage is:
Considerations

www.americanradiohistory.com

1.

162

R

-F

Tube

Vacuum

Rig

(e5)2

-

1052

-

2XPg 2X0.53

Amplifiers

As in the case of the Class B audio amplifier the grid resistance of the linear
amplifier varies from infinity to a low
value when maximum grid current is
drawn. To decrease the effect of this
resistance excursion, a swamping resistor should be placed across the grid tank
circuit. The value of the resistor should
be dropped until a shortage of driving
power begins to be noticed. For this example, a resistor of 3,000 ohms is used.
The grid circuit load for no grid current
is now 3,000 ohms instead of infinity,

and drops to 2400 ohms when maximum
grid current is drawn.

3.

is chosen for the grid
tank. The capacitive reactance required
A

circuit

of

Q

15

is:

-=

2400
X

=

160 ohms

15

4. At 4.0 Mc. the effective capacity is:
106

=

C=

248 µµEd.

6.28x4X154
5. The inductive reactance required to reso-

nate the grid circuit at 4.0 Mc. is:
160

L=

=

6.4 microhenries

6.28 x 4.0
6.

substituting the loaded grid resistance figure in the formula in the first
paragraph, the grid driving power is now
found to be approximately 2.3 watts.
By

Screen Circuit

Considerations

Special

8 -5

-

10,400 ohms
2.

THE

reference to the plate
characteristic curve of the
By

813 tube, it can be seen that
at a minimum plate potential of 500 volts, and
a maximum plate current of 0.6 ampere, the
screen current will be approximately 30 milliamperes, dropping to one or two milliamperes
in the quiescent state. It is necessary to use
a well -regulated screen supply to hold the
screen voltage at the correct potential over
this range of current excursion. The use of an
electronic regulated screen supply is recommended.

R

RADIO

-F Power

Amplifier Circuits
The r-f power amplifier discussions of Sections 8 -4 and 8 -5 have been based on the assumption that a conventional grounded- cathode
or cathode -return type of amplifier was in question. It is possible, however, as in the case of
a -f and low-level r -f amplifiers to use circuits
in which electrodes other than the cathode are
returned to ground insofar as the signal potential is concerned. Both the plate-return or
cathode -follower amplifier and the grid- return
or grounded -grid amplifier are effective in certain circuit applications as tuned r -f power
amplifiers.
Disadvantages of
Grounded -Cothode

Amplifiers

An

undesirable aspect of

the operation of cathode return r -f power amplifiers

using triode tubes is that
such amplifiers must be neutralized. Principles and methods of neutralizing r -f power amplifiers are discussed in the chapter Generation of R -F Energy. As the frequency of operation of an amplifier is increased the stage becomes more and more difficult to neutralize
due to inductance in the grid and plate leads
of the tubes and in the leads to the neutralizing capacitors. In other words the bandwidth
of neutralization decreases as the frequency
is increased. In addition the very presence of
the neutralizing capacitors adds additional
undesirable capacitive loading to the grid and
plate tank circuits of the tube or tubes. To
look at the problem in another way, an amplifier that may be perfectly neutralized at a frequency of 30 Mc. may be completely out of
neutralization at a frequency of 120 Mc. Therefore, if there are circuits in both the grid and
plate circuits which offer appreciable impedance at this high frequency it is quite possible that the stage may develop a "parasitic
oscillation" in the vicinity of 120 Mc.

This condition of restricted range neutralization of r -f
power amplifiers can be greatly alleviated through the use of a cathode return or grounded -grid r -f stage. The grounded grid amplifier has the following advantages:
Grounded -Grid

R-F Amplifiers

1.

The output capacitance of a stage is reduced to approximately one -half the value
which would be obtained if the same tube
or tubes were operated as a conventional
neutralized amplifier.

2.

The tendency toward parasitic oscillations
in such a stage is greatly reduced since
the shielding effect of the control grid be-

www.americanradiohistory.com

HANDBOOK

Amplifier

Grounded Grid

tween the filament and the plate is effective over a broad range of frequencies.
The feedback capacitance within the stage
3.
is the plate -to- cathode capacitance which
is ordinarily very much less than the gridto -plate capacitance. Hence neutralization
is ordinarily not required. If neutralization
is required the neutralizing capacitors are
very small in value and are cross connected between plates and cathodes in a
push -pull stage, or between the opposite
end of a split plate tank and the cathode
in a single -ended stage.
The disadvantages of a grounded -grid amplifier are:
1. A large amount of excitation energy is required. However, only the normal amount
of energy is lost in the grid circuit of the
amplifier tube; all additional energy over
this amount is delivered to the load circuit as useful output.
2. The cathode of a grounded -grid amplifier
stage is "hot" to r.f. This means that the
cathode must be fed through a suitable impedance from the filament supply, or the
secondary of the filament transformer must
be of the low- capacitance type and adequately insulated for the r -f voltage which
will be present.
3. A grounded -grid r -f amplifier cannot be
plate modulated 100 per cent unless the
output of the exciting stage is modulated
also. Approximately 70 per cent modulation of the exciter stage as the final stage
is being modulated 100 per cent is recommended. However, the grounded -grid r -f
amplifier is quite satisfactory as a Class
B linear r -f amplifier for single sideband
or conventional amplitude modulated waves
or as an amplifier for a straight c -w or
FM signal.

Figure 9 shows a simplified representation
of a grounded -grid triode r -f power amplifier
stage. The relationships between input and
out put power and the peak fundamental components of electrode voltages and currents are
given below the drawing. The calculation of
the complete operating conditions for a
grounded-grid amplifier stage is somewhat more
complex than that for a conventional amplifier
because the input circuit of the tube is in
series with the output circuit as far as the
load is concerned. The primary result of this
effect is, as stated before, that considerably
more power is required from the driver stage.
The normal power gain for a g -g stage is from
3 to 15 depending upon the grid circuit conditions chosen for the output stage. The higher
the grid bias and grid swing required on the

163

c-

{E4NEnr) IPr

PONEN OUTPUT TO LOAD

POWER DEL,ORSED

n

POWER PROM DO UER TO LOAD

TOTAL PONE, DELIVERED BY

,SORKD

DIVE,

EON

5Y OUTPUT TUNE

(IPI+ I4r)
E

MD NO

.!S

E4r

Z.

(PP,o..N

E4r IPr
i

E4 =IPr

o,
POOR,

EPr IP r

Irr

Err

OUTPUT TUBE

Irr

Or E4r lc

SUPPLY

I4r

O,OrE4NIt

E4r

I.N

I

lc

Figure

9

GROUNDED -GRID CLASS

B

OR

CLASS

C

AMPLIFIER
The equations in the above figure give the
relationships between the fundamental com-

ponents of grid and plate potential and current, and the power input and power output
of the stage. An expression for the approximate cathode impedance is given

output stage, the higher will be the requirement from the driver.
Calculation of Operating
Conditions of Grounded
Grid R -F Amplifiers

It is most convenient
to determine the op-

erating conditions for

a Class B or Class C
grounded -grid r -f power amplifier in a two -step
process. The first step is to determine the
plate- circuit and grid- circuit operating conditions of the tube as though it were to operate
as a conventional cathode- return amplifier
stage. The second step is then to add in the
additional conditions imposed upon the operating conditions by the fact that the stage is
to operate as a grounded -grid amplifier.
For the first step in the calculation the procedure given in Section 8 -3 is quite satisfactory and will be used in the example to follow.
Suppose we take for our example the case of a
type 304TL tube operating at 2700 plate volts
at a kilowatt input. Following through the procedure previously given:
1.

Desired power output -850 watts
Desired Plate voltage -2700 volts
Desired plate efficiency -85 per cent
(Np = 0.85)

www.americanradiohistory.com

164
2.

3.

4.

-F

R

Tube Amplifiers

Vacuum

P1° = 850/0.85

=

F,

1000 watts

850 = 150 watts
= 1000
Type 304TL chosen; max. P,
watts, /I= 12.

= 300

Ib = 1000/2700 = 0.370 ampere

(370 ma.)
5.

Approximate ipma, = 4.9 X 0.370 = 1.81
ampere

6.

epm;n= 140 volts (from 30411. con-

stant- current curves)
7.

Epm = 2700 - 140 = 2560 volts

8.

1./lb

9.

0.85 X 2700/2560 = 1.79

= 2 X

igmax/Ib

= 4.65 (from

igmax = 4.65 X 0.370 = 1.72 amperes

11.

egmp = 140 volts

12.

Cos
Op =

=

Op

= Epm /Ebb =

2560/2700 = 0.95
of 59° (from figure 6) = 0.90
Np = F, X F2 = 0.95 X 0.90 = Approx.
0.85 (85 per cent plate efficiency)
Now, to determine the operating conditions
as a grounded -grid amplifier we must also know
the peak value of the fundamental components
of plate current. This is simply equal to
(Ipm /Ib) lb, or:
Ipm = 1.79 X 0.370 = 0.660 amperes (from 4
and 8 above)
The total average power required of the
driver (from figure 9) is equal to Egmlpm /2
(since the grid is grounded and the grid swing
appears also as cathode swing) plus Pd which
is 27.5 watts from 18 above. The total is:

0.480 amperes

= 2.32

(1.79 -1.57)

=

525 X 0.660

Total drive

=

-

172.5 watts

2

figure 3)

10.

igma,

RADIO

F2 for Op

-

P,

THE

0.51

59°

plus 27.5 watts or 200 watts
Therefore the total power output of the stage
is equal to 850 watts (contributed by the
304TL) plus 172.5 watts (contributed by the
driver) or 1022.5 watts. The cathode driving
impedance of the 30411. (again referring to
figure 7) is approximately:
Zk = 525/(0.660 + 0.116) = approximately 675
ohms.

1

13.

Ecc-

-

X

1- 0.51
[0.51 (

`

_

2560

-

140)

12

-

Plate- Return or
Cathode- Follower

2700
12

J

-385 volts

14.

Egm = 140 -( -385) = 525

15.

Egm /Ecc =

16.

igmax /Ia = approx. 8.25 (extrapolated

volts

-1.36

from figure 4)

17.

la = 0.480/8.25
grid current)

18.

Pd = 0.9

19.

Pg

=

Max.

X 525 X

27.5

P,

=

0.058 (58 ma. d -c
0.058 = 27.5 watts

-( -385 X 0.058) = 5.2

watts

for 304TL is 50 watts

We can check the operating plate efficiency
of the stage by the method described in Section 8 -4 as follows:

Power Amplifier

Circuit
R

-F

diagram,

elec-

trodepotentials and currents, an d operating

conditions for a cathode- follower r -f power amplifier are given in
figure 10. This circuit can be used, in addition to the grounded -grid circuit just discussed, as an r -f amplifier with a triode tube
and no additional neutralization circuit. However, the circuit will oscillate if the impedance from cathode to ground is allowed to become capacitive rather than inductive or resistive with respect to the operating frequency. The circuit is not recommended except for
v -h -f or u -h -f work with coaxial lines as tuned
circuits since the peak grid swing required
on the r -f amplifier stage is approximately
equal to the plate voltage on the amplifier
tube if high- efficiency operation is desired.
This means, of course, that the grid tank must
be able to withstand slightly more peak voltage than the plate tank. Such a stage may not
be plate modulated unless the driver stage is
modulated the same percentage as the final
amplifier. However, such a stage may be used
as an amplifier or modulated waves (Class B
linear) or as a c -w or FM amplifier.

www.americanradiohistory.com

HANDBOOK

POWER OUTPUT TO LOAD

G

EpM

(

POWER FROM DRIVER TO LOAD

TOTAL POWER FROM DRIVER_

T.

PM. IGM)
EPM IPM
2

EPM IOM
2

(EPM +eGMP) IGM

ECU IGM
z

2

(EPM + eGMP)

AppROA

ASSUMING IGM

y

I

e

IC

c APPROA

ZG

APPROA.

I GM

O S

(Ecc

eoup) lc

( EPM+ eGMP
I

I

Figure 10
CATHODE -FOLLOWER

R

-F

Control Grid Dissipation
in Grounded -Grid Stages

Tetrode tubes maybe
operated as grounded

1.0 IC

POWER ABSORBED Or OUTPUT TUBE GRID AND BIAS SUPPLY.

-

165

circuit. If a conventional filament transformer
is to be used the cathode tank coil may consist of two parallel heavy conductors (to carry
the high filament current) by-passed at both
the ground end and at the tube socket. The
tuning capacitor is then placed between filament and ground.lt is possible in certain cases
to use two r -f chokes of special design to feed
the filament current to the tubes, with a conventional tank circuit between filament and
ground. Coaxial lines also may be used to
serve both as cathode tank and filament feed
to the tubes for v -h -f and u -h -f work.

-

POWER DELIVERED BV OUTPUT TUBE -

Amplifier

-G

POWER

AMPLIFIER
Showing the relationships between the tube
potentials and currents and the input and
output power of the stage. The approximate
grid impedance also is given.

grid

(cathode

driven) amplifiers by tying the grid and screen
together and operating the tube as a high -u
triode (figure 11). Combined grid and screen
current, however, is a function of tube geometry and may reach destructive values under
conditions of full excitation. Proper division
of excitation between grid and screen should
be as the ratio of the screen -to -grid amplification, which is approximately 5 for tubes
such as the 4 -250A, 4 -400A, etc. The proper
ratio of grid /screen excitation may be achieved by tapping the grid at some point on the
filament choke, as shown. Grid dissipation is
reduced, Lut the overall level of excitation is
increased about 30% over the value required
for simple grounded -grid operation.

The design of such an amplifier stage is
design of a
grounded -grid amplifier stage as far as the
first step is concerned. Then, for the second
step the operating conditions given in figure
10 are applied to the data obtained in the first
step. As an example, take the 304TL stage
previously described. The total power required
of the driver will be (from figure 10) approximately (2700X0.58);1.8) /2 or 141 watts. Of
this 141 watts 27.5 watts (as before) will be
lost as grid dissipation and bias loss and the
balance of 113.5 watts will appear as output.
The total output of the stage will then be approximately 963 watts.

essentially the same as the

4 -200A, 4 -400A,

fFt'

ORIVe

RFC-

RFC

rl

Cathode Tank for
G -G or C -F
Power Amplifier

The cathode tank circuit
for either a grounded-grid
or cathode -follower r -f
power amplifier may be a
conventional tank circuit if the filament transformer for the stage is of the low-capacitance
high- voltage type. Conventional filament transformers, however, will not operate with the
high values of r -f voltage present in such a

FIGURE II
TAPPED FILAMENT CHORE REDUCES EXCESSIVE
GRID DISSIPATION IN G -G CIRCUIT.

RFC--TWOP
CAC.,
TAP

71- e.a

II

WINDINGS OP 191C WIRE, ES TURN!
DIAM. TOTAL LENGTH IS SIR INCHES. GRID
TURNS PROM GROUND CND OP ONE WINDING.

I-

1

vOLTS AT
AMPERE!. (VOLTAGE DROP ACROSS
I.a VOLTS )

RFC Is

www.americanradiohistory.com

-F

Vacuum

Tube

THE

,'Amplifiers

166

R

8 -6

Class ABI Radio Frequency
Power Amplifiers

»
yll
i
iiiiiis
vis
w
w1i
/j
o

PORTIONOF

EG-tP
Class Aß1 r -f amplifiers operate under
such conditions of bias and excitation that
grid current does not flow over any portion of
the input cycle. This is desirable, since
distortion caused by grid current loading is
absent, and also because the stage is capable
of high power gain. Stage efficiency is about

i 91JJiT1-!ßr
-

MOSTLINRWA,-P
CURVE

.Ì*n

15111111111114

MIN

RADIO

A,

SIGNAL

111

IN

plate current operating angle of
2100 is chosen, as compared to 62% for Class
58% when a

operation.
The level of static (quiescent) plate current
for lowest distortion is quite critical for
Class AB1 tetrode operation. This value is
determined by the tube characteristics, and is
not greatly affected by the circuit parameters
or operating voltages. The maximum d -c plate
potential is therefore limited by the static
dissipation of the tube, since the resting plate
current figure is fixed. The static plate current
of a tetrode tube varies as the 3/2 power of
the screen voltage. For example, raising the
screen voltage from 300 to 500 volts will
double the plate current. The optimum static
plate current for minimum distortion is also
doubled, since the shape of the Eg -Ip curve
does not change.
In actual practice, somewhat lower static
plate current than optimum may be employed
without raising the distortion appreciably,
and values of static plate current of 0.6 to
0.8 of optimum may be safely used, depending
upon the amount of nonlinearity that can be
tolerated.
As with the class B linear stage, the minimum plate voltage swing of the class AB1
amplifier must be kept above the d-c screen
potential to prevent operation in the nonlinear
portion of the characteristic curve. A low value
of screen voltage allows greater r -f plate
voltage swing, resulting in improvement in
plate efficiency of the tube. A balance between plate dissipation, plate efficiency, and
plate voltage swing must be achieved for best
linearity of the amplifier.
B

The 5 -Curve

The perfect linear amplifier delivers a signal
that is a replica of the input signal. Inspection
of the plate characteristic curve of a typical
tube will disclose the tube linearity under
class A operating conditions (figure 12). The
curve is usually of exponential shape, and
the signal distortion is held to a small value
by operating the tube well below its maximum
output, and centering operation over the most
linear portion of the characteristic curve.

Figure 12
CURVE

Eg -Ip

Amplifier
most

operation is confined to
linear portion of characteristic
curve.

The relationship between exciting voltage
in a Class ABI amplifier and the r -f plate
circuit voltage is shown in figure 13. With a
small value of static plate current the lower
portion of the line is curved. Maximum undistorted output is limited by the point on the
line (A) where the instantaneous plate voltage
down to the screen voltage. This "hook" in
the line is caused by current diverted from
the plate to the grid and screen elements of
the tube. The characteristic plot of the usual
linear amplifier takes the shape of an S-curve.
The lower portion of the curve is straightened
out by using the proper value of static plate
current, and the upper portion of the curve is
avoided by limiting minimum plate voltage
swing to a point substantially above the value
of the

screen voltage.

The approximate operacing parameters may be
Linear Amplifier
obtained from the Constant Current curves
(Eg -Ep) or the Eg -Ip curves of the tube in
question. An operating load line is first
approximated. One end of the load line is
determined by the d -c operating voltage of the
tube, and the required static plate current.
As a starting point, let the product of the plate
voltage and current equal the plate dissipation
of the tube. Assuming we have a 4 -400A
tetrode, this end of the load line will fall on
point A (figure 14). Plate power dissipation
is 360 watts (3000V @ 120 ma). The opposite
end of the load line will fall on a point determined by the minimum instantaneous plate voltage, and by the maximum instantaneous plate
current. The minimum plate voltage, for best
linearity should be considerably higher than
Operating Parameters
for the Class ABI

www.americanradiohistory.com

HANDBOOK

ABI

R.F.

Power Amplifiers

167

therefore the load line cannot cross the Eg =0
line. At the point Ep =600, Eg =0, the maximum
plate current is 580 ma (Point "B ").
Each point the load line crosses a grid voltage axis may be taken as a point for construction of the Eg -Ip curve, just as was done in
figure 22, chapter 6. A constructed curve
shows that the approximate static bias voltage
is -74 volts, which checks closely with point A
of figure 14. In actual practice, the bias voltage is set to hold the actual dissipation slightly
below the maximum figure of the tube.

R F.
E

our

R.F.

E

iN

The single tone power output is:

Figure 13
LINEARITY CURVE OF
TYPICAL TETRODE AMPLIFIER

Emax -Eminx Ipmax. or
4

3000 -600 x .58=348 watts
4

The plate current -angle efficiency factor for
this class of operation is 0.73, and the actual

At point "A" the instantaneous plate
voltage is swinging down to the value
of screen voltage. At point "B" it is
swinging well below the screen and is
approaching the point where saturation,
or plate current limiting takes place.

plate circuit efficiency is:
Np =Emax -Emin x0.73, or
Emax

3000- 600 X 0.73 = 58.4%
3000

The power input to the stage is therefore
the screen voltage. In this case, the screen
voltage is 500, so the minimum plate voltage
excursion should be limited to 600 volts.
Class AB1 operation implies no grid current,

Pox

100 or, 348

Np

58.4

=

595 watts

The plate dissipation is: 595 -348 =247 watts.

.

4k1RtIoN

Pi- network tetrode

TOP VIEW OF A 4 -250A AMPLIFIER
amplifier may be operated Class AB, Class 8, or Class

C by varying potentials
applied to tube. Same general physical and mechanical design applies in each cose.

www.americanradiohistory.com

168

r,

RADIO

THE

-F Vacuum Tube Amplifiers

R

oo.

4 -400A rues
E SCR =

900 VOLTS

r 'S

r I0'

4

t5L-

t
POINT B

w

o

is

SO

00

oC

woo

2000

,000

S

-

ER
.s

.E

.4

m,_

LOAD LINE

.2

POINT

A

o

I

zs

SO

I
VALUE OF
ER MIN OOOV..
11.2 0.3E A

VALUE OF

MAX.
DISSIPATION
(3000 V. X 0.71

A

.300 wArrs)

75
2 nni

hiclre

14

OPERATING PARAMETERS FOR TETRODE LINEAR
AMPLIFIER ARE OBTAINED FROM CONSTANT- CURRENT
CURVES.
It can be seen that the limiting factor for
this class of operation is the static plate dissipation, which is quite a bit higher than the
operating dissipation level. It is possible, at
the expense of a higher level of distortion, to
drop the static plate dissipation and to increase
the screen voltage to obtain greater power output. If the screen voltage is set at 800, and
the bias increased sufficiently to drop the
static plate current to 90 ma, the single tone
d-c plate current may rise to 300 ma, for a
power input of 900 watts. The plate circuit
efficiency is 55.6%, and the power output is
500 watts. Static plate dissipation is 270 watts.
At a screen potential of 500 volts, the maximum screen current is less than 1 ma, and under
certain loading conditions may be negative.
When the screen potential is raised to 800 volts
maximum screen current is 18 ma. The performance of the tube depends upon the voltage
fields set up within the tube by the cathode,
control grid, screen grid, and plate. The quantity
of current flowing in the screen circuit is only
incidental to the fact that the screen is maintained at a positive potential with respect to
the electron stream surrounding it.
The tube will perform as expected so long as
the screen current, in either direction, does

not create undesirable changes in the screen
voltage, or cause excessive screen dissipation.
Good regulation of the screen supply is there-

required. Screen dissipation is highly
responsive to plate loading conditions, and the
plate circuit should always be adjusted so as
to keep the screen current below the maximum
dissipation level as established by the applied
voltage.
G -G Class B Linear
Certain tetrode and pentode
tubes, such as the 6AG7,
Tetrode Amplifier
837, and 803 perform well
as grounded grid class B linear amplifiers. In
this configuration both grids and the suppressor
are grounded, and excitation is applied to the
cathode circuit of the tube. So connected, the
tubes take on characteristics of high -mu triodes.
No bias or screen supplies are required for
this type of operation, and reasonably linear
fore

6AG7

-r

DRIVER

B4-300-700

Figure

15

SIMPLE GROUNDED -GRID

www.americanradiohistory.com

LINEAR AMPLIFIER

v.

169

HANDBOOK
V2

VI

837

837

500

803

500

803

500
250

250

250

250

VS

V4

V3

837

INPUT
2 War TS
PEAK

.001
2

N

=

.001

5KV

=

Ti
E0

115V

Figure
3

16

-STAGE KILOWATT LINEAR
AMPLIFIER FOR 80 OR 40
METER OPERATION

operation can be had with a very minimum of
circuit components (figure 15). The input impedance of the g -g stage falls between 100
and 250 ohms, eliminating the necessity of
swamping resistors, even though considerable
power is drawn by the cathode circuit of the
g -g stage.
Power gain of a g -g stage varies from approximately 20 when tubes of the 6AG7 type
are used, down to five or six for the 837 and
803 tubes. One or more g -g stages may be
cascaded to provide up to a kilowatt of power,
as illustrated in figure 16.
The input and output circuits of cascaded
g -g stages are in series, and a variation in
load impedance of the output stage reflects
back as a proportional change on the input
circuit. If the first g -g stage is driven by a high
impedance source, such as a tetrode amplifier,
any change in gain will automatically be compensated for. If the gain of V4 -V5 drops, the
input impedance to that stage will rise. This
change will reflect through V2 -V3 so that the
load impedance of VI rises. Since V1 has a
high internal impedance the output voltage
will rise when the load impedance rises. The
increased output voltage will raise the output
voltage of each g -g stage so that the overall
output is nearly up to the initial value before
the drop in gain of V4 -V5.
The tank circuits, therefore, of all g -g stages
must be resonated with low plate voltage and
excitation applied to the tubes. Tuning of one
stage will affect the ocher stages, and the input and coupling of each stage must be adjusted in turn until the proper power limit is
reached.
Operating Dota for
4 -400A Grounded
Grid Linear

Amplifier

ACH

ti

2500

An open frame filament transformer may
be used for TI. Cathode taps are ad-

justed for proper excitation of following
stage.

using a 4 -400A tube for the h.f. region. The
operating characteristics of the amplifier are
summarized in figure 17. It can be noted that
unusually low screen voltage is used on the
tube. The use of lower screen voltage has the
adverse effect of increasing the driving power,
but at the same time the static plate current
of the stage is decreased and linearity is imimproved. For grounded grid operation of the
4 -400A, a screen voltage of 300 volts (filament
to screen) gives a reasonable compromise between these factors.

OPERATING DATA FOR 4- 400A/4 -250A
G -G. LINEAR AB, AMPLIFIER
(SINGLE CONE)
D

-C SCREEN VOLTAGE

D -C

PLATE VOLTAGE

+300

+3000

+3500

60 MA.

STATIC PLATE CURRENT
D -C

+300

-60

GRID BIAS

67

PEAK CATHODE SWING

V.
V.

60 MA.

-59 V.
113 V.

MINIMUM PLATE VOLTAGE

660

MAXIMUM SIGNAL GRID CURRENT

3.6 MA.

10

MAXIMUM SIGNAL SCREEN CURRENT

.1 MA.

20 MA.

MAXIMUM SIGNAL PLATE CURRENT

195 MA.

267 MA.

MAXIMUM SIGNAL PLATE DISSIPATION

235

235

STATIC PLATE DISSIPATION

160 W.

210W.

GRID DRIVING POWER

0.63

3.4

FEEDTHRU POWER

6.55 W.

,s.e

V.

W.

W.

S00 V.

MA.

W.

W.
W.

POWER OUTPUT

(MAXIMUM)

350

W.

700

W.

POWER INPUT

(MAX /MUM)

565

W.

935

W.

Experiments have been
conducted by Collins Radio Co. on a grounded grid
linear amplifier stage

www.americanradiohistory.com

Figure

17

CHAPTER NINE

The Oscilloscope

The cathode -ray oscilloscope (also called
oscillograph) is an instrument which permits
visual examination of various electrical phenomena of interest to the electronic engineer.
Instantaneous changes in voltage, current and
phase are observable if they take place slowly enough for the eye to follow, or if they are
periodic for a long enough time so that the eye
can obtain an impression from the screen of
the cathode -ray tube. In addition, the cathode ray oscilloscope may be used to study any
variable (within the limits of its frequency
response characteristic) which can be converted into electrical potentials. This conversion is made possible by the use of some type
of transducer, such as a vibration pickup unit,
pressure pickup unit, photoelectric cell, microphone, or a variable impedance. The use
of such a transducer makes the oscilloscope
a valuable tool in fields other than electronics.

the recipient of signals from two sources: the
vertical and horizontal amplifiers. The operation of the cathode -ray tube itself has been
covered in Chapter 4; the auxiliary circuits
pertaining to the cathode -ray tube will be
covered here.
The Vertical

The incoming signal which is
to be examined is applied to
the terminals marked Vertical
Input and Ground. The Vertical Input terminal
is connected through capacitor CE (figure 2) so
that the a-c component of the input signal appears across the vertical amplifier gain control potentiometer, R,. Thus the magnitude of
the incoming signal may be controlled to provide the desired deflection on the screen of

Amplifier

To

'ERTICAL
NF T

9 -1

A

Typical Cathode -Ray
Oscilloscope

FLIP.

INTENSITY
MOD.

ATE

ro HORIZONTAL

- -So --

DEFLECTION

EAT.
T,

RE

GND

For the purpose of analysis, the operation
of a simple oscilloscope will be described.
The Du Mont type 274-A unit is a fit instrument for such a description. The block diagram
of the 274 -A is shown in figure 1. The electron beam of the cathode -ray tube can be moved

O

-- -rsa - - - -

/

NoNIZONTAL/ SWEEPS
IUT
/

R-R}DIR7
111111000111111

AM.

vertically or horizontally, or the vertical and

-

DIR
1NFVT

Figure
BLOCK DIAGRAM, TYPE 274 -A
CATHODE -RAY OSCILLOSCOPE

horizontal movements may be combined to produce composite patterns on the tube screen.
As shown in figure 1, the cathode -ray tube is

1

170
www.americanradiohistory.com

SI

STNC.

Time

Base

Generator

171

CI

0.25 LF
INPUT

6AC7

O--{

OUTPUT

VERT. AMP.

Ra

CONTROL

11A

RT.15aK

R30
ee M

Cz

R2

5100

1K

Figure

2

TYPICAL AMPLIFIER SCHEMATIC

Figure

the cathode -ray tube. Also, as shown in figure
1, S, has been incorporated to by -pass the
vertical amplifier and capacitively couple the
input signal directly to the vertical deflection

plate

if

so

desired.

In figure 2, V, is a 6AC7 pentode tube which
is used as the vertical amplifier. As the signal variations appear on the grid of
variations in the plate current of V, will take place.
Thus signal variations will appear in opposite
phase and greatly amplified across the plate
resistor, R,. Capacitor C, has been added across R, in the cathode circuit of V, to flatten
the frequency response of the amplifier at the
high frequencies. This capacitor because of
its low value has very little effect at low input frequencies, but operates more effectively
as the frequency of the signal increases. The
amplified signal delivered by V, is now applied through the second half of switch S, and
capacitor C, to the free vertical deflection
plate of the cathode -ray tube (figure 3).

V

The circuit of the horizontal
amplifier and the circuit of
the vertical amplifier, described in the above paragraph, are similar. A
switch in the input circuit makes provision for
the input from the Horizontal Input terminals
to be capacitively coupled to the grid of the
horizontal amplifier or to the free horizontal
deflection plate thus by- passing the amplifier,
or for the output of the sweep generator to be
capacitively coupled to the amplifier, as shown
in figure 1.
The Horizontal

Amplifier

The Time Bose
Generator

e l e c t r i c al
wave forms by the use of a
cathode -ray tube frequently

3

SCHEMATIC OF CATHODE -RAY TUBE
CIRCUITS
A 5BPIA cathode -ray tube is used in this
instrument. As shown, the necessary potentials for operating this tube are obtained
from a voltage divider mode up of resistors
R21 through R26 inclusive. The intensity of
the beam is adjusted by moving the contact
on R21. This adjusts the potential on the
cathode more or less negative with respect
to the grid which is operated at the full negative voltage -1200 volts. Focusing to the
desired sharpness is accomplished by adjusting the contact on R23 to provide the
correct potential for anode no. 1. Interdependency between the focus and the, intensity controls is inherent in all electrostatically focused cathode-ray tubes. In short,
there is an optimum setting of the focus control for every setting of the intensity control. The second anode of the 5BP IA is operated at ground potential in this instrument.
Also one of each pair of deflection plates is
operated at ground potential.
The cathode is operated at a high negative
potential (approximately 1200 volts) so that
the total overall accelerating voltage of this
tube is regarded os 1200 volts since the second anode is operated at ground potential.
The vertical and horizontal positioning controls which are connected to their respective
deflection plates are capable of supplying
either a positive or negative d -c potential
to the deflection plates. This permits the
spot to be positioned at any desired place
on the entire screen.

Investigation of

requires that some means be readily available
to determine the variation in these wave forms
with respect to time. When such a time base
is required, the patterns presented on the cathode -ray tube screen show the variation in amplitude of the input signal with respect to

time. Such an arrangement is made possible
by the inclusion in the oscilloscope of a Time
Base- Generator. The function of this generator is to move the spot across the screen at a
constant rate from left to right between two
selected points, to return the spot almost instantaneously to its original position, and to

www.americanradiohistory.com

172

The

Oscilloscope
884 tube, the tube

THE

RADIO

will ionize

(or fire) at a

specific plate voltage.
Capacitors C3.-C24 are selectively connected
in parallel with the 884 tube. Resistor
limits the peak current drain of the gas triode.
The plate voltage on this tube is obtained
through resistors R22,
and R11. The voltage
applied to the plate of the 884 tube cannot reach
the power supply voltage because of the charging effect this voltage has upon the capacitor
which is connected across the tube. This capacitor charges until the plate voltage becomes
high enough to ionize the gas in the tube. At
this time, the 884 tube starts to conduct and
the capacitor discharges through the tube until
its voltage falls to the extinction potential of
the tube. When the tube stops conducting, the
capacitor voltage builds up until the tube fires
again. As this action continues, it results in
the sawtooth wave form of figure 4 appearing
at the junction of
and R77.

R

Figure
SAWTOOTH

R

4

WAVE FORM

repeat this procedure at a specified rate. This
action is accomplished by the voltage output
from the time base (sweep) generator. The
rate at which this voltage repeats the cycle
of sweeping the spot across the screen is referred to as the sweep frequency. The sweep
voltage necessary to produce the motion described above must be of a sawtooth waveform, such as that shown in figure 4.
The sweep occurs as the voltage varies
from A to B, and the return trace as the voltage varies from B to C. If A-B is a straight
line, the sweep generated by this voltage will
be linear. It should be realized that the sawtooth sweep signal is only used to plot variations in the vertical axis signal with respect
to time. Specialized studies have made necessary the use of sweep signals of various
shapes which are introduced from an external
source through the Horizontal Input terminals.

R

Synchronization

the sweep generator may be
synchronized from the vertical amplifier or
from an external source. The switch S, shown
in figure 5 is mounted on the front panel to be
easily accessible to the operator.
If no synchronizing voltage is applied, the
discharge tube will begin to conduct when the
plate potential reaches the value of F.t (Firing
Potential). When this breakdown takes place
and the tube begins to conduct, the capacitor
is discharged rapidly through the tube, and the
plate voltage decreases until it reaches the
extinction potential E1. At this point conduction ceases, and the plate potential rises slowly as the capacitor begins to charge through
R7, and R25. The plate potential will again
reach a point of conduction and the circuit
will start a new cycle. The rapidity of the
plate voltage rise is dependent upon the circuit
constants R77 R25, and the capacitor selected,

The sawtooth voltage necessary to obtain the linear time
base is generated by the circuit of figure 5, which operates as follows:
A type 884 gas triode (V3) is used for the
sweep generator tube. This tube contains an
inert gas which ionizes when the voltage between the cathode and the plate reaches a certain value. The ionizing voltage depends upon
the bias voltage of the tube, which is determined by the voltage divider resistors R12 -R17.
With a specific negative bias applied to the
The Sawtooth
Generator

-

C10,0

5

ur.

GÉNECN1T011

CII,O.IUr
C 12, 03
C

T

OUTPUT

ur

1001212!

CN,22012121

SIGNAL TO

DEFLECTION PLATE

Rr
ISO

470
RE

6AC7

E

Provision has been made so

0
nTERr.A

Ree
Su

EAT.

FINE MM
CONTROL

SYNC
R a100A

1200

Figure

SCHEMATIC

OF

TO

St

5

SWEEP

GENERATOR

www.americanradiohistory.com

HANDBOOK

The

Oscilloscope

173

Eb +

EPVSEg

EP+

STATIC CONTROL

f

CHARACTERISTIC

Er

- --

Irl

I

FREE RUN
HING PERIOD

t+

D.C.GRID

BIAS

FIRING POTENTIAL
(D.C. BIAS)

Eex
FIRING

011

POTENTIA
WITH SYNC.

SIGNAL

SYNCHRONIZE
PERIOD

EXTINCTION
POTENTIAL

-EJ

SYNC. SIGNAL

APPLIED TO GRID

Figure 6
ANALYSIS OF SYNCHRONIZATION OF

C10 -C14, as well as the supply voltage

The exact relationship is given by:
Ec

Eb(1_erct

Eb

TIME -BASE

GENERATOR

time the plate potential rises to a sufficient
value, so that the sweep recurs at the same
or an integral sub-multiple of the synchronizing signal rate. This is illustrated in figure 6.

)

Figure - shows the power supply to be made up of two definite sections: a low voltage positive supply
which provides power for operating the amplifiers, the sweep generator, and the positioning
circuits of the cathode -ray tube; and the high
voltage negative supply which provides the
potentials necessary for operating the various
Power Supply

Where E,----=Capacitor voltage at time t
Eb =Supply voltage (B+ supply - cathode

bias)
Er---Firing potential or potential at which
time-base gas triode fires
Ex =Extinction potential or potential at
which time-base gas triode ceases
to conduct
e= Base of natural logarithms
t= Time in seconds
r =Resistance

c =Capacity in

in ohms

(R=T + Rzs)

farads (C10, II,

12,

,,, or

14)

The frequency of oscillation will be approximately:
1

f=rc Et-Ex
Ebl

Under this condition (no synchronizing signal applied) the oscillator is said to be /ree
running.
When a positive synchronizing voltage is
applied to the grid, the firing potential of the
tube is reduced. The tube therefore ionizes at
a lower plate potential than when no grid signal is applied. Thus the applied snychronizing voltage fires the gas -filled triode each

Figure 7
SCHEMATIC OF POWER SUPPLY

www.americanradiohistory.com

174

The Oscilloscope

THE

RADIO

I

0001

ono

o

p

o
.^

.52.40)

IÓ

.rIN

410

o

co

>

U

\r

>

Y

o

}

7

t

N
xn

¢
0
H

r

=
JV'

úu

ÿJ

Uà
VIQ
I

1\

á

ñl

ai
U

nf

z

o

I

www.americanradiohistory.com

HANDBOOK

TIME

Display

of

Waveforms

175

-+

4 SEC.
IFigure

9

PROJECTION

DRAWING OF A SINEWAVE
APPLIED TO THE VERTICAL AXIS AND A
SAWTOOTH WAVE OF THE SAME FRE-

QUENCY APPLIED SIMULTANEOUSLY
THE HORIZONTAL AXIS

ON

electrodes of the cathode -ray tube, and for
certain positioning controls.
The positive low voltage supply consists
of full -wave rectifier (V,), the output of which
is filtered by a capacitor input filter (20 -20 µfd.
and 8 II). It furnishes approximately 400 volts.
The high voltage power supply employs a half
wave rectifier tube, V,. The output of this rectifier is filtered by a resistance -capacitor filter consisting of 0.5 -0.5 pfd. and .18 M. A
voltage divider network attached from the output of this filter obtains the proper operating
potentials for the various electrodes of the
cathode -ray tube. The complete schematic of
the Du Mont 274 -A Oscilloscope is shown in
figure 8.

9 -2

Display of Waveforms

Together with a working knowledge of the
controls of the oscilloscope, an understanding
of how the patterns are traced on the screen
must be obtained for a thorough knowledge of
oscilloscope operation. With this in mind a
careful analysis of two fundamental waveform
patterns is discussed under the following
headings:
a. Patterns plotted against time (using the
sweep generator for horizontal deflection).
b. Lissajous Figures (using a sine wave for
horizontal deflection).
Patterns Plotted
Against Time

A sine wave is typical of
such a pattern and is con-

venient for this study. This

Figure 10
PROJECTION DRAWING SHOWING THE RESULTANT PATTERN WHEN THE FREQUENCY OF THE SAWTOOTH IS ONE -HALF
OF THAT EMPLOYED IN FIGURE 9

amplified by the vertical amplifier
and impressed on the vertical (Y -axis) deflec-

wave is

tion plates of the cathode -ray tube. Simultaneously the sawtooth wave from the time base
generator is amplified and impressed on the
horizontal (X -axis) deflection plates.
The electron beam moves in accordance
with the resultant of the sine and sawtooth
signals. The effect is shown in figure 9 where
the sine and sawtooth waves are graphically
represented on time and voltage axes. Points
on the two waves that occur simultaneously
are numbered similarly. For example, point 2
on the sine wave and point 2 on the sawtooth
wave occur at the same instant. Therefore the
position of the beam at instant 2 is the resultant of the voltages on the horizontal and vertical deflection plates at instant 2. Referring
to figure 9, by projecting lines from the two
point 2 positions, the position of the electron
beam at instant 2 can be located. If projections were drawn from every other instantaneous position of each wave to intersect on the
circle representing the tube screen, the intersections of similarly timed projections would
trace out a sine wave.
In summation, figure 9 illustrates the principles involved in producing a sine wave trace
on the screen of a cathode -ray tube. Each intersection of similarly timed projections represents the position of the electron beam acting under the influence of the varying voltage
waveforms on each pair of deflection plates.
Figure 10 shows the effect on the pattern of
decreasing the frequency of the sawtooth

www.americanradiohistory.com

176

The

Oscilloscope

THE

RADIO

B

Figure

12

METHOD OF CALCULATING FREQUENCY
RATIO OF LISSAJOUS FIGURES

Figure 11
PROJECTION DRAWING SHOWING THE RE-SULTANT LISSAJOUS PATTERN WHEN A
SINE WAVE APPLIED TO THE HORIZONTAL AXIS IS THREE TIMES THAT APPLIED TO THE VERTICAL AXIS

wave. Any recurrent waveform plotted against
time can be displayed and analyzed by the
same procedure as used in these examples.
The sine wave problem just illustrated is
typical of the method by which any waveform
can be displayed on the screen of the cathode ray tube. Such waveforms as square wave,
sawtooth wave, and many more irregular recurrent waveforms can be observed by the same
method explained in the preceding paragraphs.

9 -3

Obtaining a Lissalous
Pattern on the screen
Oscilloscope Settings

1.

The horizontal am-

plifier should be discon-

nected from the sweep
oscillator. The signal
to be examined should be connected to the
horizontal amplifier of the oscilloscope.
2. An audio oscillator signal should be connected to the vertical amplifier of the oscilloscope.
3. By adjusting the frequency of the audio
oscillator a stationary pattern should be obtained on the screen of the oscilloscope. It is
not necessary to stop the pattern, but merely
to slow it up enough to count the loops at the
side of the pattern.
4. Count the number of loops which intersect
an imaginary vertical line AB and the number
of loops which intersect the imaginary horizontal line BC as in figure 12. The ratio of
the number of loops which intersect AB is to

Lissajous Figures

Another fundamental pattern is the Lissajous
figure, named after the 19th century French
scientist. This type of pattern is of particular
use in determining the frequency ratio between
two sine wave signals. If one of these signals
is known, the other can be easily calculated
from the pattern made by the two signals upon
the screen of the cathode -ray tube. Common
practice is to connect the known signal to the
horizontal channel and the unknown signal to
the vertical channel.
The presentation of Lissajous figures can
be analyzed by the same method as previously
used for sine wave presentation. A simple example is shown in figure 11. The frequency
ratio of the signal on the horizontal axis to the
signal on the vertical axis is 3 to 1. If the
known signal on the horizontal axis is 60 cycles per second, the signal on the vertical
axis is 20 cycles.

O

RATIO

I

O

I

RATIO

O RATIO 5
Figure 13
OTHER LISSAJOUS PATTERNS

www.americanradiohistory.com

2I

HANDBOOK

HHASE

Lissajous

DIFFERENCE =O

PHASE DIFFERENCE

190

PHASE DIFFERENCE

PHASE

-5

OIFFERENCE'225

Figure
LISSAJOUS PATTERNS OBTAINED

PHASE DIFFERENCE

Figure 13 shows other examples of Lissa jous figures. In each case the frequency ratio
shown is the frequency ratio of the signal on
the horizontal axis to that on the vertical
ve r ti c
axis.
Phase Differonce Patterns

Coming under the heading of
Lissajous figures is the method
used to determine the phase
difference between signals of the same frequency. The patterns i n vol v e d take on the
form of ellipses with different degrees of ec-

centricity.
The following steps should

be taken to obphase -difference pattern:
1. With no signal input to the oscilloscope,
the spot should be centered on the screen
of the tube.
2. Connect one signal to the vertical amplifier of the oscilloscope, and the other
signal to the horizontal amplifier.
3. Connect a common ground between the
two frequencies under investigation and
the oscilloscope.

.90.

270

177

PHASE DIFFERENCE=135

PHASE DIFFERENCE

315

14

FROM THE

the number of loops which intersect BC as the
frequency of the horizontal signal is to the
frequency of the vertical signal.

tain

PHASE DIFFERENCE

Figures

MAJOR PHASE

DIFFERENCE ANGLES

plifier control is adjusted

(3 inches). Reconnect the signal to the vertical amplifier.
The resulting pattern will give an accurate
picture of the exact phase difference between
the two waves. If these two patterns are exactly the same frequency but different in phase
and maintain that difference, the pattern on
the screen will remain stationary. If, 'however,
one of these frequencies is drifting slightly,
the pattern will drift slowly through 360°. The
phase angles of 0 °, 45 °, 90 °, 135 °, 180 °,
225 °, 270 °, 315° are shown in figure 14.
Each of the eight patterns in figure 14 can
be analyzed separately by the previously used

a

Adjust the vertical amplifier gain so as
to give about 3 inches of deflection on a
5 inch tube, and adjust the calibrate d
scale of the oscilloscope so that the vertical axis of the scale coincides precisely with the vertical deflection of the spot.
5. Remove the signal from the vertical amplifier, being careful not to change the
setting of the vertical gain control.
6. Increase the gain of the horizontal amplifier to give a deflection exactly the
same as that to which the vertical am-

TIME

-

4.

Figure 15
PROJECTION DRAWING SHOWING THE RESULTANT PHASE DIFFERENCE PATTERN
OF TWO SINE WAVES 45° OUT OF PHASE

www.americanradiohistory.com

178

Oscilloscope

The

Y

Y

INTERCEPT =O

/

SINE

Y

THE

MAXIMUM =I
Y

MAXIMUM.

SINE=

MAXIMUM

=

I

'I MA IMU

MAXIMUM= I

SINE e=

_S

So

INTERCEPT'.S

SINEe',
s=150-

projecti ',n method. Figure 15 shows two sine
waves which differ in phase being projected
on to the screen of the cathode -ray tube. These
signals represent a phase difference of 45 °.
It is extremely important: (1) that the spot
has been centered on the screen of the cathode ray tube, (2) that both the horizontal and vertical amplifiers have been adjusted to give
exactly the same gain, and (3) that the calibrated scale be originally set to coincide with
the displacement of the signal along the vertical axis. If the amplifiers of the oscilloscope
are not used for conveying the signal to the
deflection plates of the cathode -ray tube, the
coarse frequency switch should be set to horizontal input direct and the vertical input

MODULATED

Y

'I

DIFFERENCE

Figure 17
MODULATION

I

¿;-:.

Figure 16
EXAMPLES SHOWING THE USE OF THE FORMULA

TRAPEZOIDAL

=

INTERCEPT =.5

s' o

YINTERCEil,
s=so

Y

RADIO

FOR DETERMINATION OF

PHASE

switch to direct and the outputs of the two
signals must be adjusted to result in exactly
the same vertical deflection as horizontal deflection. Once this deflection has been set by
either the oscillator output controls or the amplifier gain controls in the oscillograph, it
should not be changed for the duration of the
measurement.

Determination of
the Phase Angle

The relation commonly used
in determining the phase
angle between signals is:
Y intercept

Sine

9

Y maximum

PATTERN

Figure 18
CARRIER WAVE PATTERN

Figure 19
PROJECTION DRAWING SHOWING TRAPEZOIDAL PATTERN

www.americanradiohistory.com

HANDBOOK

Trapezoidal Pattern

MODULATED
CARRIER

R F.

179

POWER AMPLIFIER

1-0

ANTENNA

TIME

EACH 1M,

1

MODULATOR 500.11ÁF

STAGE

-

/SAW TOOTH

10000 V.
TV CAPACITOR

SWEEP

CRO
LC TUNES

TO OPERATING FREQUENCY

Figure 20
PROJECTION DRAWING SHOWING MODOLATED CARRIER WAVE PATTERN

C

e+

intercept

=
=

Y maximum

=

where
Y

9

phase angle between signals
point where ellipse crosses vertical axis measured in tenths of
inches. (Calibrations on the
calibrated screen)
highest vertical point on ellipse
in tenths of inches

Several examples of the use of the formula are
given in figure 16. In each case the Y intercept and Y maximum are indicated together
with the sine of the angle and the angle itself.
For the operator to observe these various patterns with a single signal source such as the
test signal, there are many types of phase
shifters which can be used. Circuits can be
obtained from a number of radio text books.
The procedure is to connect the original signal to the horizontal channel of the oscilloscope and the signal which has passed through
the phase shifter to the vertical channel of
the oscilloscope, and follow the procedure set
forth in this discussion to observe the various
phase shift patterns.

9 -4

Monitoring Transmitter
Performance with the Oscilloscope

The oscilloscope may be used as an aid for
the proper operation of a radiotelephone transmitter, and may be used as an indicator of the
overall performance of the transmitter output
signal, and as a modulation monitor.

There are two types of patterns
that can serve as indicators, the
trapezoidal pattern (figure 17) and the modu-

Waveforms

NOTE'

IF

L

_

PICKUP IS INSUFFICIENT,
A TUNED CIRCUIT MAY BE USED
AT THE OSCILLOSCOPE AS SHOWN.
R F.

Figure 21
MONITORING CIRCUIT FOR TRAPEZOIDAL MODULATION PATTERN

laced wave pattern (figure 18). The trapezoidal
pattern is presented on the screen by impressing a modulated carrier wave signal on the vertical deflection plates and the s i g n a l t h a t
modulates the carrier wave signal (the modulating signal) on the horizontal deflection
plates. The trapezoidal pattern can be analyzed by the method used previously in analyzing waveforms. Figure 19 shows how the signals cause the electron beam to trace out the

pattern.
The modulated wave pattern is accomplished
by presenting a modulated carrier wave on the
vertical deflection plates and by using the
time -base generator for horizontal deflection.
The modulated wave pattern also can be used
for analyzing waveforms. Figure 20 shows how
the two signals cause the electron beam to
trace out the pattern.

oscilloscope connections for obtaining a trapezoidal pattern are shown in
figure 21. A portion of the audio output of the
transmitter modulator is applied to the horizontal input of the oscilloscope. The vertical
amplifier of the oscilloscope is disconnected,
and a small amount of modulated r -f energy is
coupled directly to the vertical d e f l e c t i o n
plates of the oscilloscope. A small pickup
loop, loosely coupled to the final amplifier
tank circuit and connected to the vertical deThe Trapezoidal
Pattern

www.americanradiohistory.com

The

Oscilloscope

The

180

T

H E

RADIO

i

T

EMIN

E

MAX

1
TRAPEZOIDAL WAVE PATTERN
Figure

22

Figure

(L ESS THAN 100^; MODULATION)

(100

mula:

Emax
Emax

=

- Emin x
t

Emin

Figure 24

MODULATION)

flection plates by a short length of coaxial
line will suffice. The amount of excitation to
the plates of the oscilloscope may be adjusted
to provide a pattern of convenient size. Upon
modulation of the transmitter, the trapezoidal
pattern will appear. By changing the degree of
modulation of the carrier wave the shape of
the pattern will change. Figures 22 and 23
show the trapezoidal pattern for various degrees of modulation. The percentage of modulation may be determined by the following forModulation percentage

23

(OVER MODULATION)

figure 25. The internal sweep circuit of the
oscilloscope is applied to the horizontal
plates, and the modulated r -f signal is applied
to the vertical plates, as described before. If
desired, the internal sweep circuit may be snychronized with the modulating signal of the
transmitter by applying a small portion of the
modulator output signal to the external sync
post of the oscilloscope. The percentage of
modulation may be determined in the same
fashion as with a trapezoidal pattern. Figures
26, 27 and 28 show the modulated wave pattern for various degrees of modulation.

100

where Emax and Emin are defined as in
figure 22.
An overmodulated signal is shown in figure

9 -5

Receiver -F Alignment
with an Oscilloscope
I

24.
The Modulated
Wove Pattern

R

F.

The oscilloscope connections
for obtaining a modulated
wave pattern are shown in

POWER AMPLIFIER

TO

CRO

ANTENNA
USE INTERNAL
SWEE

a
FROM
MODUL ATOR
LC TUNES TO OP-

ERATING FREQUENCY

Figure 25
MONITORING CIRCUIT FOR
MODULATED WAVE PATTERN

The alignment of the i -f amplifiers of a receiver consists of adjusting all the tuned circuits to resonance at the intermediate frequency and at the same time to permit passage of
a predetermined number of side bands. The
best indication of this adjustment is a resonance curve representing the response of the
i -f circuit to its particular range of frequencies.
As a rule medium and low- priced receivers
use i -f transformers whose bandwidth is about
5 kc. on each side of the fundamental frequency. The response curve of these i -f transformers is shown in figure 29. High fidelity receivers usually contain i -f transformers which
have a broader bandwidth which is usually 10
kc. on each side of the fundamental. The response curve for this type transformer is shown
in figure 30.

Resonance curves such as these can be displayed on the screen of an oscilloscope. For
a complete understanding of the procedure it
is important to know how the resonance curve
is traced.

www.americanradiohistory.com

HANDBOOK

Receiver Alignment

VV
EMIN

E

181

V\I"

MLY.

1\1\
CARRIER WAVE PATTERN

Figure 27

Figure 26

(100% MODULATION)

(LESS THAN 100% MODULATION)

The Resonance
Curve on the
Screen

To present a resonance curve
on the screen, a frequency modulated signal source must
be

avail a b l

e.

Figure 28

This signal

source is a signal generator whose output is
the fundamental i -f frequency which is frequency- modulated 5 to 10 kc. each side of the
fundamental frequency. A signal generator of
this type generally takes the form of an ordinary signal generator with a rotating motor
driven tuned circuit capacitor, called a uwob-

(OVER MODULATION)

bulator, or its electronic equivalent,

a

react-

ance tube.
The method of presenting a resonance curve
on the screen is to connect the vertical channel of the oscilloscope across the detector
load of the receiver as shown in the detectors
of figure 31 (between point A and ground) and
the time -base generator output to the horizontal channel. In this way the d -c voltage across
the detector load varies with the frequencies
which are passed by the i -f system. Thus, if
the time -base generator is set at the frequency
of rotation of the motor driven capacitor, or
the reactance tube, a pattern resembling figure 32, a double resonance curve, appears on
the screen.

Figure 32 is explained by considering fighalf a rotation of the motor driven
capacitor the frequency increases from 445
kc. to 465 kc., more than covering the range
of frequencies passed by the i -f system.
Therefore, a full resonance curve is presented
on the screen during this half cycle of rotation since only half a cycle of the voltage producing horizontal deflection has transpired.
In the second half of the rotation the motor
ure 33. In

eKC

K
4

KC ecc

Figure 29
FREQUENCY RESPONSE CURVE OF THE
I -F OF A LOW PRICED RECEIVER

ecc

TRIODE DETECTOR

eKc

DIODE DETECTOR

Figure

Figure 30
FREQUENCY RESPONSE OF
HIGH- FIDELITY I -F SYSTEM

31

CONNECTION OF THE OSCILLOSCOPE
ACROSS THE DETECTOR LOAD

www.americanradiohistory.com

The

182

Figure 32
DOUBLE RESONANCE CURVE

445

KC

455 KC

THE

Oscilloscope

46

455 KC

KC

445 KC

Figure 33
DOUBLE
RESONANCE ACHIEVED BY
COMPLETE ROTATION OF THE MOTOR
DRIVEN CAPACITOR

curve is observed as it sweeps the spot across
the screen from left to right; and it is observed
again as the sine wave sweeps the spot back
again from right to left. Under these conditions the two response curves are superimposed on each other and the high frequency
responses of both curves are at one end and
the low frequency response of both curves is
at the other end. The i -f trimmer capacitors
are adjusted to produce a response curve
which is symmetrical on each side of the fundamental frequency.
When using sawtooth sweep, the two response curves can also be superimposed. If
the sawtooth signal is generated at exactly
twice the frequency of rotation of the motor
driven capacitor, the two resonance curves
will be superimposed (figure 34) if the i -f
transformers are properly tuned. If the two
curves do not coincide the i -f trimmer capacitors should be adjusted. At the point of coincidence the tuning is correct. It should be
pointed out that rarely do the two curves agree
perfectly. As a result, optimum adjustment is
made by making the peaks coincide. This latter procedure is the one generally used in i -f
adjustment. When the two curves coincide, it
is evident that the i -f system responds equally to signals higher and lower than the fundamental i -f frequency.
9 -6

Figure

34

SUPER -POSITION OF RESONANCE CURVES

driven capacitor takes the frequency of the
signal in the reverse order through the range
of frequencies passed by the i -f system. In
this interval the time -base generator sawtooth
waveform completes its cycle, drawing the
electron beam further across the screen and
then returning it to the starting point. Subsequent cycles of the motor driven capacitor and
the sawtooth voltage merely retrace the same
pattern. Since the signal being viewed is applied through the vertical amplifier, the sweep
can be synchronized internally.
Some signal generators, particularly those
employing a reactance tube, provide a sweep
output in the form of a sine wave which is
synchronized to the frequency with which the
reactance tube is swinging the fundamental
frequency through its limits, usually 60 cycles
per second. If such a signal is used for horizontal deflection, it is already synchronized.
Since this signal is a sine wave, the response

RADIO

Single Sideband Applications

Measurement of power output and distortion
are of particular importance in SSB transmitter
adjustment. These measurements are related to
the extent that distortion rises rapidly when
the power amplifier is overloaded. The useable
power output of a SSB transmitter is often defined as the maximum peak envelope power

II11IIIIIIU114ulul

m

IIIIIIIIIIIIIIIIIIII

Figure
SINGLE

35

TONE PRESENTATION

Oscilloscope trace of SSB signal
modulated by single tone (A).
Incomplete carrier supression or
spurious products will show
modulated envelope of (B). The
ratio of supression is:

www.americanradiohistory.com

S

-

20 log

A

+B

A -B

S.S.B.

HANDBOOK

R

POWER
VER A

-F INPUT

T

R -F

LIFIER

SSB INPUT
VOLTAGE

FOM

TEST

GERMANIUM
DIODE

tpplications
2.5

183

MM

RFC

AUDIO OUTPUT

70 OSCILLOSCOPE

DIVIDER OR
PICMUP COIL

INPUT

ENVELOPE
DETECTOR

Figure 37
SCHEMATIC OF
ENVELOPE DETECTOR

OSCILLOSCOPE

Figure 36
BLOCK DIAGRAM OF
LINEARITY TRACER

obtainable with a specified signal-to- distortion
ratio. The oscilloscope is a useful instrument
for measuring and studying distortion of all
types that may be generated in single sideband
equipment.
Single Tone

When aSSB

transmitter is modu-

laced with a single audio tone,
the r -f output should be a single
radio frequency. If the vertical plates of the
oscilloscope are coupled to the output of the
transmitter, and the horizontal amplifier sweep
is set to a slow rate, the scope presentation
will be as shown in figure 35. If unwanted distortion products or carrier are present, the top
and bottom of the pattern will develop a "ripple" proportional to the degree of spurious
Observations

products.
The linearity tracer is an auxiliary detector to be used with
an oscilloscope for quick observation of amplifier adjustments and parameter variations. This instrument consists of
two SSB envelope detectors the outputs of
which connect to the horizontal and vertical
inputs of an oscilloscope. Figure 36 shows a
block diagram of atypical linearity test set -up.
A two -tone test signal is normally employed
to supply a SSB modulation envelope, but any
modulating signal that provides an envelope
that varies from zero to full amplitude may be
The Linearity

Tracer

used. Speech modulation gives a satisfactory
trace, so that this instrument may be used as
a visual monitor of transmitter linearity. It is
particularly useful for monitoring the signal
level and clearly shows when the amplifier
under observation is overloaded. The linearity
trace will be a straight line regardless of the
envelope shape if the amplifier has no distortion. Overloading causes a sharp break in
the linearity curve. Distortion due to too much
bias is also easily observed and the adjustment
for low distortion can easily be made.
Another feature of the linearity detector is

that the distortion of each individual stage
can be observed. This is helpful in troubleshooting. By connecting the input envelope
detector to the output of the SSB generator,
the overall distortion of the entire r -f circuit
beyond this point is observed. The unit can
also serve as a voltage indicator which is
useful in making tuning adjustments.
The circuit of a typical envelope detector
is shown in figure 37. Two matched germainum
diodes are used as detectors. The detectors
are not linear at low signal levels, but if the
nonlinearity of the two detectors is matched,
the effect of their nonlinearity on the oscilloscope trace is cancelled. The effect of diode
differences is minimized by using a diode load
of 5,000 to 10,000 ohms, as shown. It is important that both detectors operate at approximately the same signal level so that their
differences will cancel more exactly. The
operating level should be 1 -volt or higher.
It is convenient to build the detector in a
small shielded enclosure such as an i -f transformer can fitted with coaxial input and output
connectors. Voltage dividers can be similarly
constructed so that it is easy to insert the desired amount of voltage attenuation from the
various sources. In some cases it is convenient
to use a pickup loop on the end of a short
length of coaxial cable.
The phase shift of the amplifiers in the oscilloscope should be the same and their frequency response should be flat out to at least
twenty times the frequency difference of the
two test tones. Excellent high frequency characteristics are necessary because the rectified
SSB envelope contains harmonics extending
to the limit of the envelope detector's response.
Inadequate frequency response of the vertical
amplifier may cause a little "foot" to appear
on the lower end of the trace, as shown in
figure 38. If it is small, it may be safely neg-

lected.
Another spurious effect often encountered
is a double trace, as shown in figure 39. This
can usually be corrected with an R -C network
placed between one detector and the oscilloscope. The best method of testing the detectors
and the amplifiers is to connect the input of

www.americanradiohistory.com

184

The

Oscilloscope

OUTPUT

SIGNAL
LEVEL

Figure 38
EFFECT OF INADEQUATE
RESPONSE
OF
VERTICAL
AMPLIFIER
INPUT SIGNAL LEVEL

Figure 41
ORDINATES ON LINEARITY
CURVE
FOR
3RD
ORDER
DISTORTION EQUATION

Figure 39
DOUBLE TRACE
CAUSED BY PHASE
SHIFT

the envelope detectors in parallel. A perfectly
straight line trace will result when everything
is working properly. One detector is then connected to the other r -f source through a voltage
divider adjusted so that no appreciable change
in the setting of the oscilloscope amplifier
controls is required. Figure 40 illustrates some

typical linearity traces. Trace A is caused by
inadequate static plate current in class A or
class B amplifiers or a mixer stage. To regain
linearity, the grid bias of the stage should be
reduced, the screen voltage should be raised,
or the signal level should be decreased. Trace
B is a result of poor grid circuit regulation
when grid current is drawn, or a result of non-

linear plate characteristics of the amplifier
tube at large plate swings. More grid swamping
should be used, or the exciting signal should
be reduced. A combination of the effects of A
and B are shown in Trace C. Trace D illustrates
amplifier overloading. The exciting signal
should be reduced.
A means of estimating the distortion level
observed is quite useful. The first and third
order distortion components may be derived by
an equation that will give the approximate
signal -to- distortion level ratio of a two tone
test signal, operating on a given linearity curve.
Figure 41 shows a linearity curve with two
ordinates erected at half and full peak input
signal level. The length of the ordinates et
and e2 may be scaled and used in the following
equation:
Signal -to- distortion ratio in db =20 log 8 e t -e2

TYPICAL LINEARITY TRACES

Figure 40

TYPICAL LINEARITY
TRACES

www.americanradiohistory.com

2

el -e2

CHAPTER TEN

Special Vacuum Tube Circuits

A whole new concept of vacuum tube applications has been developed in recent years.
No longer are vacuum tubes chained to the
field of communication. This chapter is devoted to some of the more common circuits encountered in industrial and military applications of the vacuum tube.

10 -1

The characteristics of a
diode tube are such that the
tube conducts only when the plate is at a positive potential with respect to the cathode. A
positive potential may be placed on the cathode, but the tube will not conduct until the
voltage on the plate rises above an equally
positive value. As the plate becomes more
positive with respect to the cathode, the diode
conducts and passes that portion of the wave
that is more positive than the cathode voltage.
Diodes may be used as either series or parallel limiters, as shown in figure 1. A diode may
be so biased that only a certain portion of the
positive or negative cycle is removed.
Diode Limiters

Limiting Circuits

The term limiting refers to the removal or
suppression by electronic means of the extremities of an electronic signal. Circuits
which perform this function are referred to as
limiters or clippers. Limiters are useful in
wave-shaping circuits where it is desirable to
square off the extremities of the applied signal. A sine wave may be applied to a limiter
circuit to produce a rectangular wave. A
peaked wave may be applied to a limiter circuit to eliminate either the positive or negative peaks from the output. Limiter circuits
are employed in FM receivers where it is necessary to limit the amplitude of the signal applied to the detector. Limiters may be used to
reduce automobile ignition noise in short -wave
receivers, or to maintain a high average level
of modulation in a transmitter. They may also
be used as protective devices to limit input
signals to special circùits.

An audio peak clipper consisting
of two diode limiters may be used
to limit the amplitude of an audio signal to a predetermined value to provide
a high average level of modulation without
danger of overmodulation. An effective limiter
for this service is the series -diode gate clipper. A circuit of this clipper is shown in figure 2. The audio signal to be clipped is coupled to the clipper through C,. R, and R2 are
the clipper input and output load resistors.
The clipper plates are tied together and are
connected to the clipping level control, R.,
through the series resistor, R3. R. acts as a
voltage divider between the high voltage supply and ground. The exact point at which clipAudio Peak

Limiting

185
www.americanradiohistory.com

Special Vacuum Tube

186

e iN

Circuits

THE

RADIO

e OUT
E

IAA'
E

e

IN

=

VOLTAGE DROP
ACROSS DIODE

E= VOLTAGE DROP
ACROSS DIODE

e OUT

PIN

e OVT

E

E

lTV i

VT

-A-21

ear

Figure

e OUT

1

VARIOUS DIODE LIMITING CIRCUITS
Series diodes limiting positive and negative peaks are shown in A and
ing positive and negative peaks are shown in C and D. Parallel diodes 8. Parallel diodes limitlimiting above and below
ground are shown in E and F. Parallel diode limiters which pass
negative and positive peaks
are shown in G and H.

ping will occur is set by R,, which controls the
positive potential applied to the diode plates.
Under static conditions, a d -c voltage is obtained from R4 and applied through R, to both
plates of the 6AL5 tube. Current flows through
R,,
and divides through the two diode
sections of the 6AL5 and the two load resistors, R, and Rr. All parts of the clipper circuit
are maintained at a positive potential above
ground. The voltage drop between the plate
and cathode of each diode is very small compared to the drop across the 300,000 -ohm resistor (R,) in series with the diode plates.
The plate and cathode of each diode are therefore maintained at approximately equal potentials as long as there is plate current flow.
Clipping does not occur until the peak audio
input voltage reaches a value greater than the
static voltages at the plates of the diode.

R

Assume that R4 has been set to a point that
will give 4 volts at the plates of the 6AL5.
When the peak audio input voltage is less than
4 volts, both halves of the tube conduct at all
times. As long as the tube conducts, its resistance is very low compared with the plate
resistor R,. Whenever a voltage change occurs
across input resistor
the voltage at all of
the tube elements increases or decreases by
the same amount as the input voltage change,
and the voltage drop across R, changes by an
equal amount. As long as the peak input voltage is less than 4 volts, the 6AL5 acts merely
as a conductor, and the output cathode is permitted to follow all voltage changes at the input cathode.
If, under static conditions, 4 volts appear at
the diode plates, then twice this voltage (8
volts) will appear if one of the diode circuits

www.americanradiohistory.com

R

Clamping Circuits

HANDBOOK
Ra

6AL5

CLIPPING
LEVEL

300K

CONTROL

C2

Ci
0.1

0.1

e

e IN

OUT

R
00

m

R2

R
zoom

eIN

E

Bt

E

200K

Figure

2

THE SERIES -DIODE GATE CLIPPER FOR
AUDIO PEAK LIMITING

is opened, removing its d -c load from the circuit. As long as only one of the diodes continues to conduct, the voltage at the diode
plates cannot rise above twice the voltage selected by R. In this example, the voltage cannot rise above 8 volts. Now, if the input audio
voltage applied through C, is increased to any
peak value between zero and plus 4 volts, the
first cathode of the 6AL5 will increase in voltage by the same amount to the proper value between 4 and 8 volts. The other tube elements
will assume the same potential as the first
cathode. However, the 6AL5 plates cannot increase more than 4 volts above their original
4 -volt static level. When the input voltage to
the first cathode of the 6AL5 increases to
more than plus 4 volts, the cathode potential
increases to more than 8 volts. Since the plate
circuit potential remains at 8 volts, the first
diode section ceases to conduct until the input voltage across R, drops below 4 volts.
When the input voltage swings in a negative
direction, it will subtract from the 4 -volt drop
across R, and decrease the voltage on the input cathode by an amount equal to the input
voltage. The plates and the output cathode will
follow the voltage level at the input cathode
as long as the input voltage does not swing
below minus 4 volts. If the input voltage does
not change more than 4 volts in a negative
direction, the plates of the 6AL5 will also become negative. The potential at the output
cathode will follow the input cathode voltage
and decrease from its normal value of 4 volts
until it reaches zero potential. As the input
cathode voltage decreases to less than zero,
e

IVENPOSITIVE
A
WNEDNGRIDOSEDR

Figure 3
LIMITING CIRCUIT

GRID

the plates will follow. however, the output
will stop at zero
cathode, grounded through
potential as the plate becomes negative. Conduction through the second diode is impossible
under these conditions. The output cathode
remains at zero potential until the voltage at
the input cathode swings back to zero.
The voltage developed across output resistor R2 follows the input voltage variations as
long as the input voltage does not swing to a
peak value greater than the static voltage at
the diode plates, determined by R. Effective
clipping may thus be obtained at any desired

R

level.

The square-topped audio waves generated
this clipper are high in harmonic content,
but these higher order harmonics may be greatby

ly reduced by a low -level speech filter.
Grid Limiters

A

triode grid limiter is shown

in figure 3. On

positive peaks

of the input signal, the triode grid attempts to
swing positive, and the grid- cathode resistance drops to a value on the order of 1000
ohms or so. The voltage drop across R (usually of the order of I megohm) is large compared to the grid-cathode drop, and the resulting limiting action removes the top part of the
positive input wave.

Clamping Circuits

10 -2

A circuit which holds either amplitude extreme of a waveform to a given reference level

e OUT

IN

187

eiN

eouT

DIODE CONDUCTS

OA POSITIVE CLAMPING CIRCUIT

SIMPLE POSITIVE

©

NEGATIVE CLAMPING CIRCUIT

Figure 4
AND NEGATIVE CLAMPING CIRCUITS

www.americanradiohistory.com

188

e

Special Vacuum Tube Circuits
r

-,

L-

J

RADIO

THE

¡DEFLECTION
COIL

IN

-100Y
CI CHARGE PATH

NEGATIVE
PLOYED IN

Figure 5
CLAMPING CIRCUIT EMELECTROMAGNETIC SWEEP

C2 DISCHARGE PATH

Figure 7
THE CHARGE AND DISCHARGE PATHS
IN FREE -RUNNING MULTIVIBRATOR OF
FIGURE 6

SYSTEM

is repeated and therefore is "jittery." If a
clamping circuit is placed between the sweep
amplifier and the deflection element, the start
of the sweep can be regulated by adjusting the
d -c voltage applied to the clamping tube (fig-

B+

ure 5).

Multivibrators

10-3
Figure 6
BASIC MULTIVIBRATOR CIRCUIT

The multivibrator, or relaxation oscillator,
is used for the generation of nonsinusoidal
waveforms. The output is rich in harmonics,
but the inherent frequency stability is poor.
The multivibrator may be stabilized by the
introduction of synchronizing voltages of harmonic or subharmonic frequency.
In its simplest form, the multivibrator is a
simple two -stage resistance -capacitance coupled amplifier with the output of the second
stage coupled through a capacitor to the grid
of the first tube, as shown in figure 6. Since
the output of the second stage is of the proper
polarity to reinforce the input signal applied
to the first tube, oscillations can readily take
place, started by thermal agitation noise and

of potential is called a clamping circuit or a
d-c restorer. Clamping circuits are used after
RC cpupling circuits where the wave f o r m
swing is required to be either above or below
the reference voltage, instead of alternating
on both sides of it (figure 4). Clamping circuits are usually encountered in oscilloscope
sweep circuits. If the sweep voltage does not
always start from the same reference point,
the trace on the screen does not begin at the
same point on the screen each time the sweep

B.

B.

NIP
///

SYNCHNONIZING

SIGNAL

B

DIRECT- COUPLED CATHODE
MULTI VIBRATOR

ELECTRON-COUPLED

MULTIVIBRATOR

Figure

©

MULTI VIBRATOR WITH SINE -WAVE
SYNCHRONIZING SIGNAL APPLIED
TO ONE TUBE

8

VARIOUS FORMS OF MULTIVIBRATOR CIRCUITS

www.americanradiohistory.com

-

Multivibrators

189

PULSE
OUTPUT

ONE -SHOT MULTIVIBRATOR

BASIC ECCLES-JORDAN TRIGGER

CIRCUIT

Figure

9

ECCLES -JORDAN MULTI VIBRATOR CIRCUITS

)

miscellaneous tube noise. Oscillation is maintained by the process of building up and discharging the store of energy in the grid coupling capacitors of the two tubes. The charging and discharging paths are shown in figure
7. Various forms of multivibrators are shown
in figure 8.
The output of a multivibrator may be used
as a source of square waves, as an electronic
switch, or as a means of obtaining frequency
division. Submultiple frequencies as low as
one -tenth of the injected synchronizing frequency may easily be obtained.
The Eccles- Jordan

Circuit

The Eccles -Jordan trigger
circuit is shown in figure
9A. This is not a true mul-

tivibrator, but rather a circuit that possesses
two conditions of stable equilibrium. One condition is when V, is conducting and V2 is cutoff; the other when V2 is conducting and V, is
cutoff. The circuit remains in one or the other

of these two stable conditions with no change
in operating potentials until some external
action occurs which causes the nonconducting
tube to conduct. The tubes then reverse their
functions and remain in the new condition as
long as no plate current flows in the cutoff
tube. This type of circuit is known as a flip -

flop circuit.

Figure 9B illustrates a modified Eccles Jordan circuit which accomplishes a complete
cycle when triggered with a positive pulse.

Such a circuit is called a one -shot multivibrator. For initial action, V, is cutoff and V2 is

conducting. A large positive pulse applied to
the grid of V, causes this tube to conduct, and
the voltage at its plate decreases by virtue of
the IR drop through R3. Capacitor C2 is charged
rapidly by this abrupt change in V, plate voltage, and V, becomes cutoff while V, conducts.
This condition exists until C2 discharges, allowing V2 to conduct, raising the cathode bias
of V, until it is once again cutoff.
A direct, cathode -coupled multivibrator is
shown in figure 8A. RK is a common cathode
resistor for the two tubes, and coupling takes
place across this resistor. It is impossible for
a tube in this circuit to completely cutoff the
other tube, and a circuit of this type is called
a free- running multivibrator in which the condition of one tube temporarily cuts off the
other.

RF
PULSE

RF

RF

PULSE

PULSE

nns

nnl

'1

eoUT

nnl
CUTOFF

TIME

C ouT

I_

Figure 10
BLOCKING

TIME

TIMG

MNI3CIC04E

CUTOFF
TIME

SINGLE -SWING

CUTOFF

OSCILLATOR

Figure

11

HARTLEY OSCILLATOR USED AS BLOCKING
OSCILLATOR BY PROPER CHOICE OF R, -C,

190

Special Vacuum

eIN

Tube

Circuits

e

Pour

POSITIVE COUNTING CIRCUIT

THE

eouT

NEGATIVE COUNTING CIRC,

POSITIVE

'.ETEA

Figure
POSITIVE AND NEGATIVE

10 -4

The Blocking Oscillator

A blocking oscillator is any oscillator which
cuts itself off after one or more cycles caused
by the accumulation of a negative charge on
the grid capacitor. This negative charge may
gradually be drained off through the grid resistor of the tube, allowing the circuit to oscillate once again. The process is repeated
and the tube becomes an intermittent oscillator. The rate of such an occurance is determined by the R-C time constant of the grid circuit. A single -swing blocking oscillator is
shown in figure 10, wherein the tube is cutoff
before the completion of one cycle. The tube
produces single pulses of energy, the time
between the pulses being regulated by the
discharge time of the grid R -C network. The
self-pulsing blocking oscillator is shown in
figure 11, and is used to produce pulses
of r-f energy, the number of pulses being de-

termined by the timing network in the grid circuit of the oscillator. The rate at which these
pulses occur is known as the pulse -repetition
frequency, or p.r. /.

10 -5

ADIO

IN

..

NG
)N

CIRCUIT WITH

12

COUNTING CIRCUITS

ing units to be counted, and produces a- voltage that is proportional to the frequency of
the pulses. A counting circuit may be used in
conjunction with a blocking oscillator to produce a trigger pulse which is a submultiple of
of the frequency of the applied pulse. Either
positive or negative pulses may be counted.
A positive counting circuit is shown in figure
12A, and a negative counting circuit is shown
in figure 12B. The positive counter allows a
certain amount of current to flow through R,
each time a pulse is applied to C,.
The positive pulse charges
and makes
the plate of V, positive with respect to its
cathode. V, conducts until the exciting pulse
passes. C, is then discharged by
and the
circuit is ready to accept another pulse. The
average current flowing through R, increases
as the pulse- repetition frequency increases,
and decreases as the p.r.f. decreases.
By reversing the diode connection s, as
shown in figure 12B, the circuit is made to

C

V

respond to negative pulses. In this circuit, an
increase in the p.r.f. causes a decrease in the
average current flowing through
which is
opposite to the effect in the positive counter.

R

Counting Circuits

A counting circuit, or frequency divider is
one which receives uniform pulses, represent-

e

Vs

feIN

3

e OUTy

Figure
Figure 13
STEP -BY -STEP COUNTING CIRCUIT

14

The step -by -step counter used to trigger a
blocking oscillator. The blocking oscillator
serves as a frequency divider.

www.americanradiohistory.com

HANDBOOK

R

-C

Oscillators

191

4

C2

CI

!00'1.

C

VI

R=LP

RB

LP

R4. 4

a1

xCI =R2

THE

WATT, 110
X

V

LAMP BULB

Ca

Figure 15
WIEN- BRIDGE AUDIO OSCILLATOR

A step- counter is similar to the circuits
discussed, except that a capacitor which is
large compared to C, replaces the diode load
resistor. The charge of this condenser is increased during the time of each pulse, producing a step voltage across the output (figure
13). A blocking oscillator may be connected
to a step- counter, as shown in figure 14. The
oscillator is triggered into operation when the
voltage across C, reaches a point sufficiently
positive to raise the grid of V, above cutoff.
Circuit parameters may be chosen so that a
count division up to 1/20 may be obtained
with reliability.

10 -6

Resistance -Capacity

Oscillators
In an R -C oscillator, the frequency is determined by a resistance capacity network that
provides regenerative coupling between the
output and input of a feedback amplifier. No
use is made of a tank circuit consisting of inductance and capacitance to control the frequency of oscillation.
The Wien - Bridge oscillator employs a Wien
network in the R -C feedback circuit and is
shown in figure 15. Tube V, is the oscillator
tube, and tube V, is an amplifier and phase inverter tube. Since the feedback voltage
through C4 produced by V, is in phase with the
input circuit of V, at all frequencies, oscillation is maintained by voltages of any frequency that exist in the circuit. The bridge circuit
is used, then, to eliminate feedback voltages
of all frequencies except the single frequency
desired at the output of the oscillator. The
bridge allows a voltage of only one frequency
to be effective in the circuit because of the
degeneration and phase shift provided by this

Figure 16
PHASE -SHIFT OSCILLATOR

THE

circuit. The frequency at which oscillation
occurs is:

f-

1
,

2n R, C,

when

R,xC,=R,xC,

Lp is used as the cathode resistor
thermal stabilizer of the oscillator
amplitude. The variation of the resistance
with respect to current of the lamp bulb holds
the oscillator output voltage at a nearly constant amplitude.
The phase -shi /t oscillator shown in figure
16 is a single tube oscillator using a three
section phase shift network. Each section of
the network produces a phase shift in proportion to the frequency of the signal that passes
through it. For oscillations to be produced,
the signal from the plate of the tube must be
shifted 180 °. Three successive phase shifts
of 60° accomplish this, and the frequency of
oscillation is determined by this phase shift A high -mu triode or a pentode must be used in
this circuit. In order to increase the frequency
of oscillation, either the resistance or the
capacitance must be decreased.
A lamp

of V, as

THE

a

Figure 17
BRIDGE -TYPE PHASE -SHIFT
OSCILLATOR

www.americanradiohistory.com

192

Special

Vacuum

Tube

Circuits

THE

RADIO

0-FREQ.

OF

OSCILLATION

NEC F/B =POS F/B

-NOTCHFREQUENCY

F-

NEGATIVE

I

2?RC

FEEDBACK

WHERE

(LOOP

C=1/Ct C2

2)

POSITIVE

FEEDBACK
(LOOP r)

f
rFREQ

OF

OSCILLATION

PHASE

SHIFT'0

"NOTCH NETWORK

Figure 19
BRIDGE -T FEEDBACK
LOOP CIRCUITS

Figure 18
THE NBS BRIDGE -T
OSCILLATOR CIRCUIT AS USED
IN THE HEATH AG -9 AUDIO
GENERATOR

A bridge -type phase shill oscillator is
shown in figure 17. The bridge is so proportioned that at only one frequency is the phase
shift through the bridge 180 °. Voltages of other
frequencies are fed back to the grid of the tube
out of phase with the existing grid signal, and
are cancelled by being amplified out of phase.
The NBS Bridge -T oscillator developed by
the National Bureau of Standards consists of
a two stage amplifier having two feedback
loops, as shown in figure 18. Loop 1 consists
of a regenerative cathode-to- cathode loop, consisting of Lp, and C3, The bulb regulates the
positive feedback, and tends to stabilize the
output of the oscillator, much as in the manner of the Wien circuit. Loop 2 consists of a
grid -cathode degenerative circuit, containing
the bridge -T. Oscillation will occur at the
null frequency of the bridge, at which frequency the bridge allows minimum degeneration
in loop 2 (figure 19).

Oscillation will occur at the null
frequency of the bridge, at which
frequency
the
bridge allows
minimum degeneration in loop 2.

effect system. The furnace (F) raises the
room temperature (T) to a predetermined value
at which point the sensing thermostat (TAI)
reduces the fuel flow to the furnace. When the
room temperature drops below the predetermined value the fuel flow is increased by the
and

thermostat control. An interdependent control
system is created by this arrangement: the
room temperature depends upon the thermostat
action, and the thermostat action depends upon
the room temperature. This sequence of events
may be termed a closed loop feedback system.

ROOM

FURNACE

TEMPERATURE

(F)

(T)

FEEDBACK
(ERROR SIGNAL)

10 -7

Feedback

discus

Feedback amplifiers have been
sed
in Chapter 6, section 15 of this Handbook. A
more general use of feedback is in automatic
control and regulating systems. Mechanical
feedback has been used for many years in such
forms as engine speed governors and steering
servo engines on ships.
A simple feedback system for temperature
control is shown in figure 20. This is a cause

Figure 20
SIMPLE CLOSED LOOP
FEEDBACK SYSTEM

Room temperature (T) controls
fuel supply to furnace (F) by feedloop through Thermostat
back
(TH) control.

www.americanradiohistory.com

Feedback

HANDBOOK

INPUT SIGNAL
¡OUTPUT SIGNAL

PHASE SHIFT
OF SYSTEM

I!

TIME
OUTPUT SIGNAL

FEEDBACK SIGNAL
NO PHASE SHIFT

.. .

I

A
FEEDBACK

SIGNAL

WITH 180
PHASE
SHIFT

TIME
Figure

21

PHASE SHIFT OF ERROR
SIGNAL MAY CAUSE OSCILLATION INCLOSED LOOP SYSTEM
To prevent oscillation, the gain
of the feedback loop must be
less than unity when the phase
shift of the system reaches 180

degrees.

Error Cancellation

A feedback

control system

is dependent upon a degree
of error in the output signal, since this error
component is used to bring about the correction. This component is called the error signal.
The error, or deviation from the desired signal

193

is passed through the feedback loop to cause
an adjustment to reduce the value of the error
signal. Care must be taken in the design of
the feedback loop to reduce over -control tendencies wherein the correction signal would
carry the sytem past the point of correct operation. Under certain circumstances the new
error signal would cause the feedback control
to overcorrect in the opposite direction, resulting in hunting or oscillation of the closed
loop system about the correct operating point.
Negative feedback control would tend to
dampout spurious system oscillation if it were
not for the time lag or phase shift in the system. If the overall phase shift is equal to one half cycle of the operating frequency of the
system the feedback will maintain a steady
state of oscillation when the circuit gain is
sufficiently high, as shown in figure 21. In
order to prevent oscillation, the gain figure of
the feedback loop must be less than unity when
the phase shift of the system reaches 180 degrees. In an ideal control system the gain of
the loop would be constant throughout the
operating range of the device, and would drop
rapidly outside the range to reduce the bandwidth of the control system to a minimum.
The time lag in a closed loop system may
reduced by using electronic circuits in place
of mechanical devices, or by the use of special
circuit elements having a phase -lead characteristic. Such devices make use of the properties of a capacitor, wherein the current leads
the voltage applied to it.
be

www.americanradiohistory.com

CHAPTER ELEVEN

Electronic Computers

Mechanical computing machines were first
produced in the seventeenth century in Europe although the simple Chinese abacus (a
digital computer) had been in use for centuries. Until the last decade only simple mechanical computers (such as adding and bookkeeping machines) were in general use.
The transformation and transmission of the
volume of information required by modern

CIIP
011,

technology requires that machines assume many
of the information processing systems formerly done by the human mind. Computing machines can perform routine operations more
quickly and more accurately than a human being, processing mathematical and logistical
data on a production line basis. The computer,
however, cannot create, but can only follow
instructions. If the instructions are in error,

0111

THE IBM
COMPUTER

AND

"MEMORY"
The

puter

"704" Comis used

32,000

with

"word"

storage
memory
unit for research
Heart
programs.
of this auxiliary
unit are small,
doughnut- shaped
iron ferrites which
store information
by means of magnetism. The unit
is the first of Ms
kind to be installed with I B M's
704 computer.
Components of the
system seen in the
foreground are
(left) card punch
and (right) cord

.j

reader. In the

center of the picture is the 704's
processing unit.

www.americanradiohistory.com

Digital Computers

195

NORTN SNORE

FARN!R

CORN

N!N

^FOX

NOTE: ALL BUTTONS NAVE

ONE NORMALLY OPEN CONTACT AND

ONE NORMALLY CLOSED CONTACT.

Figure 2
A SEQUENCE COMPUTER.
Three correct buttons will sound the buzzer.

SOUTN SNORE

Figure 1
SIMPLE PUZZLES IN LOGIC MAY BE
SOLVED BY ELECTRIC COMPUTER.
THE "FARMER AND RIVER"
COMPUTER IS SHOWN HERE.

the computer will produce a wrong answer.
Computers may be divided into two classes:
the digital and the analog. The digital computer counts, and its accuracy is limited only by
the number of significant figures provided for
in the instrument. The analog computer
measures, and its accuracy is limited by the
percentage errors of the devices used, multiplied by the range of the variables they represent.

11 -1

Digital Computers

The digital computer operates in discrete
steps. In general, the mathematical operations
are performed by combinations of additions.
Thus multiplication is performed by repeated
additions, and integration is performed by
summation. The digital computer may be
thought of as an "on -off" device operating
from signals that either exist or do not exist.
The common adding machine is a simple computer of this type. The "on -off" or "yes-no"
type of situation is well suited to switches, electrical relays, or to electronic tubes.
A simple electrical digital computer may be
used to solve the old "farmer and river" problem. The farmer must transport a hen, a bushel
of corn, and a fox across a river in a small
boat capable of carrying the farmer plus one
other article. If the farmer takes the fox in
the boat with him, the hen will eat the corn.
On the other hand, if he takes the corn, the
fox will eat the hen. The circuit for a simple
computer to solve this problem is shown in
figure 1. When the switches are moved from
"south shore" to "north shore" in the proper
sequence the warning buzzer will not sound.
An error of choice will sound the buzzer.
A second simple "digital computer" is shown
is figure 2. The problem is to find the three

proper push buttons that will sound the buzzer.
The nine buttons are mounted on a board so
that the wiring cannot be seen.
Each switch of these simple computers executes an "on -off" action. When applied to a
logical problem "yes-no" may be substituted
for this term. The computer thus can act out
a logical concept concerned with a simple
choice. An electronic switch (tube) may be
substituted for the mechanical switch to increase the speed of the computer. The early
computers, such as the ENIAC (Electronic Numerical Integrator and Calculator) employed
over 18,000 tubes for memory and registering
circuits capable of "remembering" a 10 -digit
number.
11 -2

Binary Notation

To simply and reduce the cost of the digital
computer it was necessary to modify the system
of operation so that fewer tubes were used per
bit of information. The ENIAC -type computer
requires 50 tubes to register a 5 -digit number.

O

0

0

0

O

O

O

O

O

O

O

O
O

O

O

O

O

O

O
O

O

O

O

{S.

O

O

O

O

O

O
O
O
O
O

O

O

:;oFigure

0

O

0

0

O

O

O

O

3

BINARY NOTATION MAY BE USED
FOR DIGITAL DISPLAY. BINARY
BOARD ABOVE INDICATES "73092."

www.americanradiohistory.com

THE RADIO

Electronic Computers

196

O

O

-`

:O"--

BINARY NOTATION

0

o

TUBE(S)

DIGIT
I

1

a

a

3

2+1

4

4

5

+1

7

+2+1

s

4

9

!+1

10

6+2

1/

5+2+1

12

+
+4+1

14

+4+2

15

+4+2+1

1

1

2

1,0

3

1,1

s

1,0.0
1,0,1
1.1.0

6

-

4+2

12

DECIMAL NOTATION

,

7

1.1.1

0

1,0,0,0
1,0,0,1
1,0,1,0

0

10

Figure 5
BINARY NOTATION SYSTEM
REQUIRES ONLY TWO NUMBERS,
"0" AND "1."

Figure 4

BINARY DECIMAL NOTATION. ONLY
FOUR TUBES ARE REQUIRED TO
REPRESENT DIGITS FROM
TO 15.
THE DIGIT "12" IS INDICATED
1

ABOVE.

The tubes (or their indicator lamps) can be
arranged in five columns of 10 tubes each.
From right to left the columns represent units,
tens, hundreds, thousands, etc. The bottom tube
in each column represents "zero," the second
tube represents "one," the third tube "two,"
and so on. Only one tube in each column is
excited at any given instant. If the number
73092 is to be displayed, tube number seven
in the fifth column is excited, tube number
three in the fourth column, tube number zero
in the third column, etc. as shown in figure 3.
A simpler system employs the binary decimal notation, wherein any number from one
to fifteen can be represented by four tubes.
Each of the four tubes has a numerical value
that is associated with its position in the tube
group. More than one tube of the group may
be excited at once, as illustrated in figure 4.
The values assigned to the tubes in this particular group are 1, 2, 4, and 8. Additional
tubes may be added to the group, doubling the
notation of the rube thus: 1, 2, 4, 8, 16, 32,
64, 128, 356, etc. Any numerical value lower
than the highest group number can be displayed by the correct tube combination.
A third system employs the binary notation
which makes use of a bit (binary digit) representing a single morsel of information. The
binary system has been known for over forty
centuries, and was considered a mystical revelation for ages since it employed only two sym-

bols for all numbers. Computer service usually employs "zero" and "one" as these symbols.
Decimal notation and binary notation for common numbers are shown in figure 5. The
binary notation represents 4 -digit numbers
(thousands) with .ten bits, and 7 -digit numbers (millions) with 20 bits. Only one electron tube is required to display an information
bit. The savings in components and primary
power drain of a binary -type computer over
the older ENIAC -type computer is obvious.
Figure 6 illustrates a computer board showing the binary indications from one to ten.
Digital
Computer

The digital computer is em-

ployed in a "yes -no" situation. It may be used for
routine calculations that would ordinarily require enormous man -hours of time, such as
checking stress estimates in aircraft design, or
military logistics, and problems involving the
manipulation of large masses of figures.
Uses

DECIMAL NOTATION

COMPUTER NOTATION

o

O

1

o
o

a
3

0

5

0
o

4

O

O

o

0

0

0O

O

4

7

.

o
0
0

+

10

0
=

Figure

o

OFF

6

BINARY NOTATION AS REPRESENTED
ON COMPUTER BOARD FOR NUMBERS
FROM 1 TO 10.

www.americanradiohistory.com

HANDBOOK

Analog Computers

eour=e,+e2

197

+150V.

e Our

R,

e,

eour-

R,

R, +R2

e2 R2
Ri +R2

/

R3
R, R2

\[R,+ R234-R3J

Figure 7
OF TWO VOLTAGES
BY ELECTRICAL MEANS.

SUMMATION

Analog Computers

11 -3

The analog computer represents the use of
one physical system as a model for a second
system that is usually more difficult to construct or to measure, and that obeys the equations of the same form. The term analog implies similarity of relations or properties between the two systems. The common slide-rule
is a mechanical analog computer. The speedometer in an automobile is a differential analog
computer, displaying information proportional
to the rate of change of speed of the vehicle.
The electronic analog computer employs circuits containing resistance, capacitance, and inductance arranged to behave in accordance with
analogous equations. Variables are represented
by d -c voltages which may vary with time.

Figure 8
OF TWO VOLTAGES
BY ELECTRONIC MEANS.

SUMMATION

Thus complicated problems can be solved by
d -c amplifiers and potentiometer controls in
electronic circuits performing mathematical
functions.
If a linear network is energized by two voltage sources
the voltages may be summed
as shown in figure 7. Subtraction of quantities
may be accomplished by using negative and
positive voltages. A -c voltages may be employed for certain additive circuits, and more
Addition and
Subtraction

THE

HEATHKIT
ELECTRONIC

ANALOG
DIGITAL
COMPUTER
"electronic
slide rule" simuThis

lates equations or
physical problems

electronically, sub-

stituting one phys-

ical system as a
model for a sec-

system that
usually more
or costly
to construct or
measure, and that
obeys equations of
the some form.

ond
is

difficult

www.americanradiohistory.com

eour

THE RADIO

Electronic Computers

198

R2 .
tt

=

c

e

eour

Figure 9
ELECTRONIC MULTIPLICATION
MAY BE ACCOMPLISHED BY
CALIBRATED POTENTIOMETERS,
WHEN OUTPUT VOLTAGE IS
PROPORTIONAL TO THE INPUT
VOLTAGE MULTIPLIED BY A

CONSTANT (R R1).

complex circuits employ vacuum tubes, as in
figure 8. Synchronous transformers may be
used to add expressions of angular rotation,
and circuits have been developed for adding
time delays, or pulse counting.
Multiplication

Electronic multiplication and
division may be accomplished
with the use of potentiometers where the output voltage is proportional to
the input voltage multiplied by a constant
which may be altered by changing the physical
arrangement of the potentiometer (figure 9).
Variable autotransformers may also be used to
perform multiplication.
A simple bridge may be used to obtain an
output that is the product of two inputs divided
by a third input, as shown in figure 10.
and Division

Differentiation

The time derivative of a
voltage can be expressed as
a charge on a capacitor by:
de
(1)
dt
and is shown in figure 11A. The charging current is converted into a voltage by the use
of a resistor, R. If the input to the RC circuit
is charging at a uniform rate so that the current through C and R is constant, the output
voltage e is:
i

OUTPUT'\,

=C

e

our

Figure 11
ELECTRONIC DIFFERENTIATION
The time derivative of a voltage can be

expressed as a charge on a capacitor (A).
Operational amplifier (B) employs feed
back principle for short differentiation

time.

RCd

(2)

For highest accuracy, a small RC product
should be used, permitting the maximum possible differentiation time. The output of the
differentiator may be amplified to any suitable
level.
A more accurate differentiating device
makes use of an operational amplifier. This
unit is a high gain, negative feedback d -c
amplifier (discussed in section 11 -4) with the
resistance portion of the RC product appearing
in the feedback loop of the amplifier (figure
11B). A shorter differentiation time may be
employed if the junction point between R and
C could be held at a constant potential. The
feedback amplifier shown inverts the output
signal and applies it to the RC network, hold-

ing the junction potential constant.
Integration
Integration is a process of accumulation, or summation, and
requires a device capable of storing physical
quantities. A capacitor will store an electrical
charge and will give the time integral of a
current in respect to a voltage:

eo= 1

idt

(3)

In most computers, the input signal is in
the form of a voltage, and the input charging
current of the capacitor must be taken through
a series resistance as in figure 12. If the integrating time is short the charging current is
approximately proportional to the input voltage. The charging current may be made a
true measure of the input voltage by the use

RI x R3

R2

Figure 10
ELECTRONIC MULTIPLICATION BY
BRIDGE CIRCUIT PROVIDES
OUTPUT THAT IS PRODUCT OF

TWO INPUTS DIVIDED BY A
THIRD INPUT.

Figure 12
SIMPLE
INTEGRATION
CIRCUIT
Making use of
charging current
of capacitor.

www.americanradiohistory.com

e
O

T

c

eouT

HANDBOOK

Operational Amplifier

-

The Operational

-4

11

199

Amplifier

Mathematical operations are performed by
using a high gain d -c amplifier, termed an
operational amplifier. The symbol of this unit
is a triangle, with the apex pointing in the direction of operation ( figure 15). The gain of
such an amplifier is -A, so:

+

Figure 3
"MILLER FEEDBACK" INTEGRATOR
SUITABLE FOR COMPUTER USE.

eaur

eo=-Aeg,oreg=
Figure 14
R -L NETWORK USED FOR
INTEGRATION PURPOSES.

-e

(4)

A

If -A approaches infinity, e, will be approximately zero. In practice this condition is
realized by using amplifiers having open loop
gains of 30,000 to 60,000. If ea is set at 100
volts, e, will be of the order of a few millivolts. Thus, considering eg equal to zero:

o

Figure 15
OPERATIONAL AMPLIFIER ( -A)
Mathematical operations may be performed
by any operational amplifier, usually a
stable, high-gain d -c amplifier, such as

RI

-

Rf

,oreo-

Rr
RI

(5)

eI

which may be written:

shown in Figure 16.

e0

of an operational amplifier wherein the capacitance portion of the RC product appears
in the feedback loop of the amplifier, holding
the junction point between R and C at a constant potential. A simple integrator is shown
in figure 13 employing the Miller feedback
principle. Integration is also possible with an
RL network (figure 14).

_

-

Keg, where K

=

R

I

(6 )

This amounts to multiplication by a constant
coefficient, since RI and Rr may be fixed in
value. The circuit of a typical operational amplifier is shown in figure 16.
Amplifier
Operation

Two voltages may be added by
the amplifier, as shown in figure
17. Keeping in mind that eg is

u (Rv)

e,

BIAS

-4í0V
GAIN=

-ze0 V

-A

Figure 16
HIGH GAIN OPERATIONAL AMPLIFIER, SUCH AS USED IN HEATH COMPUTER.

200

THE RADIO

Electronic Computers
RF

o
e

ea

D

o

eo

Figure 17
TWO VOLTAGES MAY BE
ADDED BY SUMMATION
AMPLIFIER.

F

essentially at zero (ground) potential:

eo= eo

or,

where

Re

=

R.
K.

K=

e

Re

+

e,+K:e

(8)

and K: =

B

(7)

e2

R3

R`

As long as e., does not exceed the input
range of the amplifier, any number of inputs
may be used:

eo=

-

Re

Rl

ei

Rf
ei +
+ Á=

- - -

+

RI
Ro

en

11

By combining the above operations in various ways, problems of many kinds may be
solved .For example, consider the mass- springdamper assembly shown in figure 19. The mass
M is connected to the spring which has an
elastic constant K. The viscous damping constant is C. The vertical displacement is y. The
sum of. the forces acting on mass M is:

f
or

eo

= -Rr+

(11)

Ro

Integration is performed by replacing the
feedback resistor Re with a capacitor Ce, as
shown in figure 18. For this circuit (with ea
approximately zero):

t

but g=Cr

en, so

dc

ei

dt

Ri

d=

Ce

do

(12)

and

R

= Cf

1

and

o

e
o

R 1Cr eidt

deo

eo

ei dt

Ri Cr

(t) =M

d$

+

C

d

+Ky

(14)
(15)

.

M

dt'

-C

dt

-Ky +f (t)
(17)

If, in the analog circuit, there is a voltage

-dt

equal to M

it can be converted to
dby passing it through an integrator circuit having an RC time constant equal to M. This resulting voltage can be passed through a second
integrator stage with unit time constant which
will haue an output volage equal to y. The

voltages representing y,
Figure 18
INTEGRATION

RI
AN

oeo
o

Performed by
Summation Amplifier
by replacing feed
back resistor with a
capacitor.

(16)

where f (t) is the applied force, or forcing
function.
The first step is to set up the analog computer circuit so as to obtain an output voltage
proportional to y for a given input voltage
proportional to f (t) Equation (16) may be
rewritten in the form:

deo

(13)
Thus:

Solving Analog
Problems

-5

(10)
en

=eft/

Figure 19
"MASS- SPRINGDAMPER" PROBLEM
MAY BE SOLVED BY
ELECTRICAL ANALOGY
WITH SIMPLE
COMPUTER.

then be summed to give

(t)

can

- Ky +

f (t)

and f

-C

do

which is the right hand side of equation (17) ,
r

and therefore equal to M

www.americanradiohistory.com

dt

.

Connecting the

HANDBOOK

Analog Problems

201

SET VOLTAGE TO Y =YO As
TIME
(INITIAL

t'0

M

dZ

-2.dr

-Cdt -Kr+f(t

dt

dt 1

DISPLACEMENT)

DISPLAY
OSCILLOSCOPE
TO

Cdt - fit)

F(t)

Figure 20
ANALOG SOLUTION FOR "MASS- SPRINGDAMPER" PROBLEM OF FIGURE 19.

output of the summing amplifier (A3) to the
input of the first as shown in figure 20 satisfies the equation.
To obtain a solution to the problem, the
initial displacement and velocity must be specified. This is done by charging the integrating
capacitors to the proper voltages. Three operational amplifiers and a summing amplifier
are required.
A second problem that may be solved by
the analog computer is the example of a freely
falling body. Disregarding air resistance, the
body will fall (due to the action of gravity)
with a constant acceleration. The equation
describing this action is:
F

=mg =m

(18)

dt'

Integration of equation (18) will give the
velocity, or

,

dt

and integration a second time

will give displacement, or y. The block diagram of a suitable computer for this problem
is shown in figure 21.
If a voltage proportional to g and hence to

d'
dY

is

introduced into the first amplifier, the

-

output of that unit will he

-H

,

or the

--R2

Ri

e,=

dt

dZr
dtZ

r

-dr
dt

eo

G

Figure 21

ANALOG COMPUTER FOR
"FREELY FALLING BODY"
PROBLEM.

velocity. That, in turn, will become y, or distance, at the output of the second amplifier.
Before the problem can be solved on the
computer it is necessary to determine the time
of solution desirable and the output amplitude
of the solution. The time of solution is determined by the RC constant of the integrating
amplifiers. If RC is set at unity, computer
time is equal to real time. The computer time
desirable is determined by the method of readout. When using an oscilloscope for read -out,
a short solution time is desirable. For a recorder, longer solution time is better.
Suppose, for example, in the problem of the
falling body, the distance of fall in 2.5 seconds
is desired. Using an RC constant of 1 would
give a solution time of 2.5 seconds. This would
be acceptable for a recorder but is slow for an
oscilloscope. A convenient time of solution for
the 'scope would be 25 milliseconds. This is
1/100 of the real time, so an RC constant of
.01 is needed. This can be obtained with C
equal to 0.1 pfd, and R equal to 100,000
ohms.
It is now necessary to choose an input voltage which will not overdrive the amplifiers.
The value of g is known to be approximately
32 ft /sec. /sec. A check indicates that if we
set g equal to 32 volts, the voltage representing the answer will exceed 100 volts. Since
the linear response of the amplifier is only
100 volts, this is undesirable. An input of 16
volts, however, should permit satisfactory operation of the amplifiers. Output voltages near
zero should also be avoided. In general, output
voltage should be about 50 volts or so, with
amplifier gains of 20 to 60 being preferable.
Thus, for this particular problem the time scale factor and amplitude -scale factor have

www.americanradiohistory.com

202

THE RADIO

Electronic Computers
A'

A RELAY

RELAY

100 K

d2r

16 V.

dt2

dr

eo= Y

Figure 22
ANALOG SOLUTION FOR
"FALLING BODY PROBLEM"
OF FIGURE 21.

Figure 24

LIMITING CIRCUIT TO SIMULATE
NON -LINEAR FUNCTIONS SUCH AS

TIME IN SECONDS
4

eo

e,

tr

dt

(c

ENCOUNTERED IN HYSTERESIS,
BACKLASH, AND FRICTION
PROBLEMS.

s)

6

16

eo

32

VOLTS

DISTANCE
N

11 -6

FEET

-24

46

1

-32

Problems are frequently encountered in
which non -linear functions must be simulated.
Non -linear potentiometers may be used to supply an unusual voltage source, or diodes may
be used as limiters in those problems in which
a function is defined differently for different
regions of the independent variable. Such a
function might be defined as follows:

64

-40

-80

)

_'98

-46
36

1'00)

112

64

28

72

,44

Non -linear Functions

Figure 23
READ -OUT SOLUTION OF "FREELY
FALLING BODY" PROBLEM.

e..
e..

e..

been chosen. The problem now looks like
figure 22.
To solve the problem, relays A and A' are
opened. The solution should now appear on
the oscilloscope as shown in figure 23. The
solution of the problem leaves the integrating
capacitors charged. It is necessary to remove
this charge before the problem can be rerun.
This is done by closing relays A and A'.

--

- K:

=
K1, e,
= et, KT,, e_,
= K2, el ,, K:

KI

(19)
(20)
(21)

where K, and K: are constants.
Various limiting circuits can be used, one of
which is shown in figure 24. This is a series
limiter circuit which is simple and does not
require special components. Commonly encountered problems requiring these or similar
limiting techniques include hysteresis, backlash, and certain types of friction.
C

NOTE: REPLACE

C WITN A I MEC. RESISTOR
FOR FUNCTION SETUP

MEG
X

OUTPUT

Y

OUTPUT

RAMP - FUNCTION
GENERATOR

620

K

P
SLOPE CONTROL

E
1

2 6AL5
BREAK
CONTROL
VOLTAGE

620K

t

SIGN CHANGING

AMPLIFIER

SUMMING
AMPLIFIER

Figure 25
SIMPLIFIED

DIAGRAM OF FUNCTIONAL GENERATOR TO APPROXIMATE NON- LINEAR
FUNCTIONS.

HANDBOOK

XI

o

Xz

Non -Linear Functions

7X3

Figure 26
TYPICAL NON -LINEAR FUNCTION

WHICH MAY BE SET UP WITH
FUNCTION GENERATOR.

The Function
Generator

function generator may be
used to approximate almost
any non -linear function. This
is done by use of straight line segments which
are combined to approximate curves such as
are found in trigonometric functions as well
as in stepped functions. In a typical generator
ten line segments are used, five in the plus -x
direction, and five in the minus -x direction.
Five 6ÁL5 double diodes are used. Each line
segment is generated by a modified bridge
circuit (figure 25). A ramp function or voltage is fed into one arm of the bridge while
the opposite arm is connected to a biased diode.
The other two arms of the bridge combine to
form the output. The voltage appearing across
one of these arms is fed through a sign- changing amplifier and then summed with the voltage appearing at the opposite arm. If the arm
of potentiometer P (the slope control) is set
in the center, the bridge will be balanced and
A

Figure 27
ELECTRONIC PACKAGE IN
DIGITAL COMPUTER.
Stylized diagram of tube package. Lines carrying negative pulses are marked by a small
circle at each end. Gates are indicated by
a semi -circle with "pins" for each input.

the output of the summing amplifier will be
zero. If, on the other hand, potentiometer P
is adjusted one way or the other from center,
the bridge will be unbalanced and the summing amplifier output will vary linearly with
respect to the input in either a positive or negative y direction, depending upon which side
of center potentiometer P is set.
The break voltage, or value of x at which a
straight line segment will begin is set by
biasing the diode to the particular voltage level
or value of x desired. The ramp function generator has either a positive or negative input
which because of the 180 degree phase shift
in the amplifier, gives a minus- or plus -x output respectively. A typical function such as
shown in figure 26 may be set up with the
function generator. The initial condition volt-

IBM's new "608," the
first completely transistorized calculator
for commercial applications, operates
without the use of a
single vacuum tube.
Transistors - -tiny germanium devices that
perform many of the
functions of conventional vacuum tubes

-make

possible 50%

reduction in computer -unit size and a
in
reduction
90%
requirements
power
over a comparable
IBM tube -model machine.

They

203

are

mounted, along with
related circuitry, on
banks of printed wiring panels in the
608.
The machine's interstorage,
or "memnal
ory," is made up of
magnetic cores -minute, doughnut -shaped objects that can
"remember" information indefinitely, and
recall it for use in
calculations in a few
millionths of a
second.

www.americanradiohistory.com

204

THE RADIO

Electronic Computers

age is set to the value of X. The break -voltage
control is increased until the output of the
summing amplifier increases abruptly, indicating the diode is conducting. The input
voltage from the initial condition power supply
is set to the value X: The slope control (P)
is now set to value Y2. A second function generator may be used to set points X.1 and X.,
using the break -voltage control and the initial
condition voltage adjustments. Points XS and
X. are finally set with a third generator. The
x- output of the function generator system may
be read on an oscilloscope, using the x- output
of the ramp- function generator amplifier as
the horizontal sweep for the oscilloscope.

Digital Circuitry

11 -7

Digital circuits dealing with "and," "or,"
and "not" situations may be excited by electrical pulses representing these logical operations.
Sorting and amplifying the pulses can be accomplished by the use of electronic packages,
such as shown in figure 27. Logical operations
may be accomplished by diode -resistor gates
operating into an amplifier stage. Negative and
positive output pulses from the amplifier are
obtained through diode output gates. The driving pulses may be obtained from a standard
oscillator, operating at or near 1 mc.
A circuit of a single digital package is shown
in figure 28. Other configurations, such as a
"flip-flop" may be used. Many such packages
"AND

can be connected in series to form operational
circuits. The input "and" and "or" gates are
biased to conduction by external voltages. The
"and" diode gate transmits a pulse only when
all the input terminals are pulsed positively,
and the "or" diode gate transmits a positive
pulse applied to any one of its input terminals.
The input pulses pass through the gates and
drive the amplifier stage, which delivers an
amplified pulse to the positive and negative
output gates, and to accompanying memory
circuits.
Memory Circuits

A memory circuit consists

of some sort of delay line
which is capable of holding an information
pulse for a period of time. The amount of
delay is proportional to the frequency of the
input signal. A "long" transmission line may
be used as a delay line with the signal being
removed from the "far" end of the line after
being delayed an interval equal to the time
of transmission along the line. Lines of this
type are constructed in the manner of a
coaxial cable, except that the inner conductor
is a long, thin coil of wire. Other memory
circuits make use of magnetostrictive or piezoelectric effects to retard the pulse. Information
may also be stored in electrostatic storage
tubes, upon magnetic recording tape, and in
ferro- magnetic cores capable of holding 10,000
bits of information.

GATES

INPUT SI

LIMITING
INPUT

DIODES

2

"OR"

GATE

INPUTSS

INPUT

*4

INPUT

*5

INPUTS

6

INPUT*

7

CLAMPING
DIODES

6AN5

POSITIVE PULSE

0

NEGATIVE PULSE

INPUT

Figure 28

TYPICAL DIGITAL PACKAGE SHOWING INPUT AND OUTPUT DIODE GATES AND
PULSE AMPLIFIER.

www.americanradiohistory.com

CHAPTER TWELVE

Radio Receiver Fundamentals

A conventional reproducing device such as

loudspeaker or a pair of earphones is incapable of receiving directly the intelligence
carried by the carrier wave of a radio transmitting station. It is necessary that an additional device, called a radio receiver, be
placed between the receiving antenna and the
loudspeaker or headphones.
Radio receivers vary widely in their complexity and basic design, depending upon the
intended application and upon economic factors. A simple radio receiver for reception of
radiotelephone signals can consist of an earphone, a silicon or germanium crystal as a
carrier rectifier or demodulator, and a length
of wire as an antenna. However, such a receiver is highly insensitive, and offers no
significant discrimination between two signals in the same portion of the spectrum.
On the other hand, a dual -diversity receiver
designed for single -sideband reception and
employing double or triple detection might
occupy several relay racks and would cost
many thousands of dollars. Ilowever, conventional communications receivers are intermediate in complexity and performance between
the two extremes. This chapter is devoted to
the principles underlying the operation of
such conventional communications receivers.

12 -1

a

Detection or
Demodulation

A detector or demodulator is a device for
removing the modulation (demodulating) or
detecting the intelligence carried by an incoming radio wave.

Figure 1 illustrates an elementary form of radiotelephony receiver employing a
diode detector. Energy from a passing radio
wave will induce a voltage in the antenna and
cause a radio- frequency current to flow from
antenna to ground through coil Lt. The alternating magnetic field set up around L, links
with the turns of L2 and causes an r -f current
to flow through the parallel -tuned circuit,
1..2-C1. %hen variable capacitor C, is adjusted
so that the tuned circuit is resonant at the
frequency of the applied signal, the r-f voltage
is maximum. This r -f voltage is applied to the
diode detector where it is rectified into a varying direct current and passed through the earphones. The variations in this current correspond to the voice modulation placed on the
signal at the transmitter. As the earphone
diaphragms vibrate back and forth in accordRadiotelephony
Demodulation

205
www.americanradiohistory.com

206

Radio Receiver Fundamentals

THE

RADIO

TRIODE

lquirlli

'Ili

O
AUDIO OUTPUT

-

GROUND

L+

L2

I-

e

+

PLATE- TICKLER REGENERATION WITH "THROTTLE'
CONDENSER REGENERATION CONTROL.

Figure
ELEMENTARY FORM OF RECEIVER
This is the basis of the "crystal set" type of
1

PENTODE

ceiver, although a vacuum diode may be used in
place of the crystal diode. The tank circuit L2 -C1
is tuned to the frequency it is desired to receive.
The bypass capacitor across the phones should
have a low reactance to the carrier frequency being received, but a high reactance to the modulation on the received signal.

ance with the pulsating current they audibly
reproduce the modulation which was placed
upon the carrier wave.
The operation of the detector circuit is
shown graphically above the detector circuit
in figure 1. The modulated carrier is shown
at A, as it is applied to the antenna. B represents the same carrier, increased in amplitude,
as it appears across the tuned circuit. In C
the varying d -c output from the detector is
seen.
Radiotelegraphy
Reception

AUDIO OUTPUT

re-

+e
-e
CATHODE-TAP REGENERATION WITH SCREEN VOLTAGE
REGENERATION CONTROL.

Figure 2
REGENERATIVE DETECTOR CIRCUITS
Regenerative detectors are seldom used at the present time due to their poor selectivity. However,
they do illustrate the simplest type of receiver
which may be used either for radiophone or radiotelegraph reception.

Since a c -w telegraphy sig-

nal consists of an unmodulated carrier which is interrupted to form dots and dashes, it is apparent
that such a signal would not be made audible
by detection alone. While the keying is a form
of modulation, it is composed of such low frequency components that the keying envelope
itself is below the audible range for hand keying speeds. Some means must be provided
whereby an audible tone is heard while the
unmodulated carrier is being received, the tone
stopping immediately when the carrier is in-

terrupted.
The most simple means of accomplishing
this is to feed a locally generated carrier of
a slightly different frequency into the same
detector, so that the incoming signal will mix
with it to form an audible beat note. The difference frequency, or heterodyne as the beat
note is known, will of course stop and start
in accordance with the incoming c -w radiotelegraph signal, because the audible heterodyne can exist only when both the incoming
and the locally generated carriers are present.

The Autodyne

Detector

The local signal which is used
to beat with the desired c -w
signal in the detector may be

supplied by a separate low-power oscillator
in the receiver itself, or the detector may be
made to self -oscillate, and thus serve the
dual purpose of detector and oscillator. A detector which self -oscillates to provide a beat
note is known as an autodyne detector, and
the process of obtaining feedback between the
detector plate and grid is called regeneration.
An autodyne detector is most sensitive when
it is barely oscillating, and for this reason
a regeneration control is always included in
the circuit to adjust the feedback to the proper
amount. The regeneration control may be either
a variable capacitor or a variable resistor,
as shown in figure 2.
With the detector regenerative but not oscillating, it is also quite sensitive. When the
circuit is adjusted to operate in this manner,
modulated signals may be received with considerably greater strength than with a non regenerative detector.

www.americanradiohistory.com

Superregenerative Detectors

HANDBOOK
12 -2

Superregenerative
Receivers

ultra -high frequencies, when it is desired to keep weight and cost at a minimum,
a special form of the regenerative receiver
known as the superregenerator is often used
for radiotelephony reception. The superregenerator is essentially a regenerative receiver
with a means provided to throw the detector
rapidly in and out of oscillation. The frequency
at which the detector is made to go in and out
of oscillation varies with the frequency to be
received, but is usually between 20,000 and
500,000 times a second. This superregenerative action considerably increases the sensitivity of the oscillating detector so that the
usual "background hiss" is greatly amplified
when no signal is being received. This hiss
diminishes in proportion to the strength of the
received signal, loud signals eliminating the
hiss entirely.

TO

At

There are two systems in common
use for causing the detector to break
in and out of oscillation rapidly. In
one, a separate interruption -frequency oscillator is arranged so as to vary the voltage rapidly on one of the detector tube elements (usually the plate, sometimes the screen) at the high
rate necessary. The interruption- frequency
oscillator commonly uses a conventional tickler- feedback circuit with coils appropriate for
its operating frequency.
The second, and simplest, type of super regenerative detector circuit is arranged so
as to produce its own interruption frequency
oscillation, without the aid of a separate tube.
The detector tube damps (or "quenches ") itself
out of signal- frequency oscillation at a high
rate by virtue of the use of a high value of
grid leak and proper size plate- blocking and
grid capacitors, in conjunction with an excess
of feedback. In this type of "self- quenched"
detector, the grid leak is quite often returned
to the positive side of the power supply (through
the coil) rather than to the cathode. A representative self-quenched superregenerative detector circuit is shown in figure 3.
Except where it is impossible to secure
sufficient regenerative feedback to permit
superregeneration, the self-quenching circuit
is to be preferred; it is simpler, is self- adjusting as regards quenching amplitude, and can
have good quenching wave form. To obtain
as good results with a separately quenched
superregenerator, very careful design is required. However, separately quenched circuits
are useful when it is possible to make a certain tube oscillate on a very high frequency
but it is impossible to obtain enough regeneration for self-quenching action.
Quench
Methods

2J7

AUDIO

AMPLIFIER

Figure 3
SUPERREGENERATIVE DETECTOR CIRCUIT
A sell -quenched superregenerative detector such
as illustrated above is capable of giving good
sensitivity in the v-h -f range. However, the circuit
has the disadvantage that its selectivity is relatively poor. Also, such o circuit should be preceded by an r -I stage to suppress the radiation of
o signal by the oscillating detector.

The optimum quenching frequency is a function of the signal frequency. As the operating
frequency goes up, so does the optimum quenching frequency. hen the quench frequency is
too low, maximum sensitivity is not obtained.
When it is too high, both sensitivity and selectivity suffer. In fact, the optimum quench frequency for an operating frequency below 15 Mc.
is in the audible range. This makes the superregenerator impracticable for use on the lower

frequencies.

The high background noise or hiss which
is heard on a properly designed superregenerator when no signal is being received is not
the quench frequency component; it is tube
and tuned circuit fluctuation noise, indicating
that the receiver is extremely sensitive.
A moderately strong signal will cause the
background noise to disappear completely,
because the superregenerator has an inherent
and instantaneous automatic volume control
characteristic. This same a -v-c characteristic
makes the receiver comparatively insensitive
to impulse noise such as ignition pulses -a
highly desirable feature. This characteristic
also results in appreciable distortion of a received radiotelephone signal, but not enough
to affect the intelligibility.
The selectivity of a superregenerator is
rather poor as compared to a superheterodyne,
but is surprisingly good for so simple a receiver when figured on a percentage basis
rather than absolute kc. bandwidth.
FM Reception

A.

superregenerative receiver

will receive frequency modulated signals with results comparing favorably
with amplitude modulation if the frequency
swing of the FM transmitter is sufficiently
high. For such reception, the receiver is detuned slightly to either side of resonance.

www.americanradiohistory.com

208

Radio

Receiver Fundamentals

THE

RADI

O

AUDIO
OUTPUT

22 MC

OUTPUT

IINTERMED.

RF

AMPLIFIER

I

"SECOND*

REOUENCV

AMPLIFIER
I_

AUDIO

'AMPLIFIER

DETECTOR

1

I

-

0+100V

FREONCY
IOSCILLATORI

(FOR C.W.)

I

AUOIO

Figure 5
ESSENTIAL UNITS OF A
SUPERHETERODYNE RECEIVER
The basic portions of the receiver are shown
in solid blocks. Practicable receivers employ the dotted blocks and also usually include such additional circuits os a noise

limiter,

Figure 4
THE FREMODYNE SUPERREGENERATIVE

SUPERHETERODYNE DETECTOR FOR
FREQUENCY MODULATED SIGNALS

Superregenerative receivers radiate a strong,
broad, and rough signal. For this reason, it is
necessary in most applications to employ a
radio frequency amplifier stage ahead of the
detector, with thorough shielding throughout
the receiver.
The Fremodyne

Detector

The Hazel tin e- Fremodyne
superregenerative circuit is
expressly designed for re-

ception of FM signals. This versatile circuit
combines the action of the superregenerative
receiver with the superhetrodyne, converting
FM signals directly into audio signals in one
double triode tube (figure 4). One section of
the triode serves as a superregenerative mixer,
producing an i -f of 22 Mc., an i -f amplifier, and
a FM detector. The detector action is accomplished by slope detection tuning on the side
of the i -f selectivity curve.
This circuit greatly reduces the radiated
signal, characteristic of the superregenerative
detector, yet provides many of the desirable
features of the superregenerator. The pass band of the Fremodyne detector is about
400 kc.

12 -3

Superheterodyne

Receivers
Because of its superiority and nearly universal use in all fields of radio reception, the

circuit, and a crystal
in the i -f amplifier.

on a -v -c

filter

theory of operation of the superheterodyne
should be familiar to every radio student and
experimenter. The following discussion concerns superheterodynes for amplitude- modulation reception. It is, however, applicable in
part to receivers for frequency modulation.

Principle of

In the superheterodyne, the incoming signal is applied to a
mixer consisting of a non -linear
impedance such as a vacuum tube or a diode.
The signal is mixed with a steady signal generated locally in an oscillator stage, with the
result that a signal bearing all the modulation
applied to the original signal but of a frequency equal to the difference between the
local oscillator and incoming signal frequencies appears in the mixer output circuit. The
output from the mixer stage is fed into a fixed tuned intermediate -frequency amplifier, where
it is amplified and detected in the usual manner, and passed on to the audio amplifier. Figure 5 shows a block diagram of the fundamental superheterodyne arrangement. The basic
components are shown in heavy lines, the
simplest superheterodyne consisting simply
of these three units. However, a good communications receiver will comprise all of the
elements shown, both heavy and dotted blocks.
Operation

Superheterodyne
Advantages

The advantages of super heterodyne reception are
directly attributable to the
use of the fixed -tuned intermediate -frequency
(i -f) amplifier. Since all signals are converted
to the intermediate frequency, this section of
the receiver may be designed for optimum selectivity and high amplification. High amplification is easily obtained in the intermediatefrequency amplifier, since it operates at a

www.americanradiohistory.com

HANDBOOK

The

advantage over the tuned
radio frequency (t -r -f) type of receiver because
of what is commonly known as arithmetical
selectivity.
This can best be illustrated by considering
two receivers, one of the t -r -f type and one of
the superheterodyne type, both attempting to

PENTODE

1

To

BY -PASS

MEG.

CAPACITORS 05 TO 0.1 JIPD.

A V C

Figure

6

TYPICAL I -F AMPLIFIER STAGE

relatively low frequency, where conventional
pentode-type tubes give adequate voltage gain.
A typical i -f amplifier is shown in figure 6.
From the diagram it may be seen that both
the grid and plate circuits are tuned. The tuned
circuits used for coupling between i -f stages
are known as i-I transformers. These will be
more fully discussed later in this chapter.
Choice of Intermediate Frequency

The choice of a frequency
for the i -f amplifier involves several considera-

tions. One of these considerations concerns
selectivity; the lower the intermediate frequency the greater the obtainable selectivity.
On the other hand, a rather high intermediate
frequency is desirable from the standpoint of
image elimination, and also for the reception
of signals from television and FM transmitters
and modulated self -controlled oscillators, all
of which occupy a rather wide band of frequencies, making a broad selectivity characteristic
desirable. Images are a pecularity common to
all superheterodyne receivers, and for this
reason they are given a detailed discussion
later in this chapter.
While intermediate frequencies as low as
50 kc. are used where extreme selectivity is
a requirement, and frequencies of 60 Mc. and
above are used in some specialized forms of
receivers, most present -day communications
superheterodynes use intermediate frequencies
around either 455 kc. or 1600 kc.
Home -type broadcast receivers almost always use an i -f in the vicinity of 455 kc.,
while auto receivers usually use a frequency
of about 262 kc. The standard frequency for
the i -f channel of FM receivers is 10.7 Mc.
Television receivers use an i -f which covers
the band between about 21.5 and 27 Mc., although a new band between 41 and 46 Mc. is
coming into more common usage.
Arithmetical

Selectivity

209

an overwhelming

VARIABLE-1/

INPIp

Superhetrodyne

Aside from allowing the use of
fixed -tuned band -pass amplifier
stages, the superheterodyne has

receive a desired signal at 10,000 kc. and
eliminate a strong interfering signal at 10,010
kc. In the t -r -f receiver, separating these two
signals in the tuning circuits is practically
impossible, since they differ in frequency by
only 0.1 per cent. However, in a superheterodyne with an intermediate frequency of, for example, 1000 kc., the desired signal will be
converted to a frequency of 1000 kc. and the
interfering signal will be converted to a frequency of 1010 kc., both signals appearing at
the input of the i -f amplifier. In this case, the
two signals may be separated much more readily, since they differ by 1 per cent, or 10 times
as much as in the first case.
The converter stage, or mixer,
of a superheterodyne receiver
can be either one of two types:
(1) it may use a single envelope converter
tube, such as a 6K8, 6SA7, or 6BE6, or (2) it
may use two tubes, or two sets of elements in
the same envelope, in an oscillator -mixer arrangement. Figure 7 shows a group of circuits
of both types to illustrate present practice
with regard to types of converter stages.
Converter tube combinations such as shown
in figures 7A and 7B are relatively simple and
inexpensive, and they do an adequate job for
most applications. With a converter tube such
as the 6SB7 -Y or the 6BA7 quite satisfactory
performance may be obtained for the reception
of relatively strong signals (as for example
FM broadcast reception) up to frequencies in
excess of 100 Mc. However, the equivalent input noise resistance of such tubes is of the
order of 200,000 ohms, which is a rather high
value indeed. So such tubes are not suited for
operation without an r -f stage in the high frequency range if weak -signal reception is
The Converter
Stage

desired.

The 6L7 mixer circuit shown in figure 7C,
and the 6BA7 circuit of figure 71), also are
characterized by an equivalent input noise re-

sistance of several hundred thousand ohms, so

that these also must be preceded by one or
more r-f stages with a fairly high gain per
stage if a low noise factor is desired of the
complete receiver.
However, the circuit arrangements shown
at figures 7F and 6F are capable of low -noise
operation within themselves, so that these
circuits may be fed directly from the antenna
without an r -f stage and still provide a good
noise factor to the complete receiver. Note

www.americanradiohistory.com

210

Radio Receiver

THE

Fundamentals

RADIO

6SÁ7, 6SB7Y,

roar
AMP

6BE6. 6BÁ7

+250

2

V

ULF

+ 50 V.

Figure

7

TYPICAL FREQUENCY- CONVERTER (MIXER) STAGES
The relative advantages of the different circuits are discussed in the text

that both these circuits use control -grid injection of both the incoming signal and the
local- oscillator voltage. Hence, paradoxically,
circuits such as these should be preceded by
an r -f stage if local- oscillator radiation is to
be held to any reasonable value of field intensity.
As the frequency of operation of
a superheterodyne receiver is increased above a few hundred megacycles the
signal -to -noise ratio appearing in the plate
circuit of the mixer tube when triodes or pentodes are employed drops to a prohibitively
low value. At frequencies above the upper -fre-

quency limit for conventional mixer stages,
mixers of the diode type are most commonly
employed. The diode may be either a vacuum tube heater diode of a special u -h -f design
such as the 9005, or it may be a crystal diode
of the general type of the 1N21 through 1N28

series.

Diode Mixers

12 -4

Mixer Noise'

and Images

The effects of mixer noise and images are
troubles common to all superheterodynes. Since

www.americanradiohistory.com

HANDBOOK

Mixer Characteristics

both these effects can largely be obviated by
the same remedy, they will be considered together.
Mixer Noise

Mixer noise of the shot- effect
type, which is evidenced by a
hiss in the audio output of the receiver, is
caused by small irregularities in the plate current in the mixer stage and will mask weak
signals. Noise of an identical nature is generated in an amplifier stage, but due to the
fact that the conductance in the mixer stage
is considerably lower than in an amplifier
stage using the same tube, the proportion of
inherent noise present in a mixer usually is
considerably greater than in an amplifier stage
using a comparable tube.
Although this noise cannot be eliminated,
its effects can be greatly minimized by placing sufficient signal- frequency amplification
having a high signal -to -noise ratio ahead of
the mixer. This remedy causes the signal output from the mixer to be large in proportion to
the noise generated in the mixer stage. Increasing the gain after the mixer will be of no
advantage in eliminating mixer noise difficulties; greater selectivity after the mixer will
help to a certain extent, but cannot be carried
too far, since this type of selectivity decreases
the i -f band -pass and if carried too far will
not pass the sidebands that are an essential
part of a voice -modulated signal.
A triode having a high trans conductance is the quietest
mixer tube, exhibiting somewhat less gain but
a better signal -to -noise ratio than a comparable multi -grid mixer tube. However, below 30
Mc. it is possible to construct a receiver that
will get down to the atmospheric noise level
without resorting to a triode mixer. The additional difficulties experienced in avoiding
pulling, undesirable feedback, etc., when using
a triode with control -grid injection tend to make
multi -grid tubes the popular choice for this
application on the lower frequencies.
On very high frequencies, where set noise
rather than atmospheric noise limits the weak
signal response, triode mixers are more widely
used. A 6J6 miniature twin triode with grids
in push -pull and plates in parallel makes an
excellent mixer up to about 600 Mc.

Triode Mixers

The amplitude of the injection volt age will affect the conversion trans conductance of the mixer, and therefore should be made optimum if maximum signal -to -noise ratio is desired. If fixed bias is
employed on the injection grid, the optimum
injection voltage is quite critical. If cathode
bias is used, the optimum voltage is not so
critical; and if grid leak bias is employed, the
Injection
Voltage

211

optimum injection voltage is not at all critical
just so it is adequate. Typical optimum injection voltages will run from 1 to 10 volts for
control grid injection, and 45 volts or so for
screen or suppressor grid injection.

There always are two signal frequencies which will combine with a given
frequency to produce the same difference frequency. For example: assume a superheterodyne with its oscillator operating on a higher
frequency than the signal, which is common
practice in present superheterodynes, tuned to
receive a signal at 14,100 kc. Assuming an
i -f amplifier frequency of 450 kc., the mixer
input circuit will be tuned to 14,100 kc., and
the oscillator to 14,100 plus 450, or 14,550 kc.
Now, a strong signal at the oscillator frequency plus the intermediate frequency (14,550
plus 450, or 15,000 kc.) will also give a difference frequency of 450 kc. in the mixer out put and will be heard also. Note that the image
is always twice the intermediate frequency
away from the desired signal. Images cause
repeat points on the tuning dial.
The only way that the image could be eliminated in this particular case would be to make
the selectivity of the mixer input circuit, and
any circuits preceding it, great enough so that
the 15,000-kc. signal never reaches the mixer
grid in sufficient amplitude to produce interference.
For any particular intermediate frequency,
image interference troubles become increasingly greater as the frequency to which the
signal- frequency portion of the receiver is
tuned is increased. This is due to the fact that
the percentage difference between the desired
frequency and the image frequency decreases
as the receiver is tuned to a higher frequency.
The ratio of strength between a signal at the
image frequency and a signal at the frequency
to which the receiver is tuned producing equal
output is known as the image ratio. The higher
this ratio, the better the receiver in regard to
image- interference troubles.
kith but a single tuned circuit between the
mixer grid and the antenna, and with 400 -500
kc. i -f amplifiers, image ratios of 60 db and
over are easily obtainable up to frequencies
around 2000 kc. Above this frequency, greater
selectivity in the mixer grid circuit through
the use of additional tuned circuits between
the mixer and the antenna is necessary if a
good image ratio is to be maintained.
Images

12 -5

Z

-F Stages

Since the necessLry tuned circuits between
the mixer and the antenna can be combined
with tubes to form r -f amplifier stages, the

www.americanradiohistory.com

Radio

212

Receiver Fundamentals
PENTODE

INPUT

6AB4,

6J6,

6J4,

12AT7

O
GROUNDED-GRID

C

RADIO

THE

-i
70

Figure

TYPICAL PENTODE

R -F

+120v.
8

AMPLIFIER STAGE
CATHODE- COUPLED

reduction of the effects of mixer noise and the
increasing of the image ratio can be accomplished in a single section of the receiver.
When incorporated in the receiver, this section is known simply as an r -/ amplifier; when
it is a separate unit with a separate tuning
control it is often known as a preselector.
Either one or two stages are commonly used
in the preselector or r -f amplifier. Some pre selectors use regeneration to obtain still
greater amplification and selectivity. An r -f
amplifier or preselector embodying more than
two stages rarely ever is employed since two
stages will ordinarily give adequate gain to
override mixer noise.

Generally speaking, atmospheric noise in the frequency
range above 30 Mc. is quite
low -so low, in fact, that the noise generated
within the receiver itself is greater than the
noise received on the antenna. Hence it is of
the greatest importance that internally generated noise be held to a minimum in a receiver.
At frequencies much above 300 Mc. there is
not too much that can be done at the present
state of the art in the direction of reducing
receiver noise below that generated in the converter stage. But in the v -h -f range, between
30 and 300 Mc., the receiver noise factor in a
well designed unit is determined by the characteristics of the first r -f stage.
The usual v -h -f receiver, whether for communications or for FM or TV reception, uses
a miniature pentode for the first r -f amplifier
stage. The 6AK5 is the best of presently available types, with the 6CB6 and the 6DC6 closely approaching the 6AK5 in performance. But
when gain in the first r -f stage is not so important, and the best noise factor must be obtained, the first r -f stage usually uses a triode.
Shown in figure 9 are four commonly used
types of triode r -f stages for use in the v -h -f
range. The circuit at (A) uses few components
and gives a moderate amount of gain with very
low noise. It is most satisfactory when the
first r -f stage is to be fed directly from a low-

6J6

+120

V

LOW NOISE

CASCODE
Lo

R -F

Stages in
the V -H -F Range

+120

V.

6BK7,6B07AOR6BZ7

200V.

Figure 9
TYPICAL TRIODE V -H -F
R -F AMPLIFIER STAGES
Triode r -f stages contribute the least amount of
noise output for a given signal level, hence their
frequent use in the v-h -f range.

impedance coaxial transmission line. Figure
9 (B) gives somewhat more gain than (A), but
requires an input matching circuit. The effective gain of this circuit is somewhat reduced
when it is being used to amplify a broad band
of frequencies since the effective Gm of the
cathode -coupled dual tube is somewhat less

www.americanradiohistory.com

HANDBOOK

The

IA MC.

Cascode

R F.

AMPLIFIER

10 MC.

TUNABLE

MIXER

213

455 KC.

MC.

TUNABLE

Amplifier

DULATOR

FII%

MIXER

I.F.

AMPLIFIER

AMPLIFIER

CRYSTAL
OSCILLATOR

VARIABLE
OSCILLATOR

AND

AUDIO

3545 KC.

1
455 KC.
MIXER

50 KC.

I I

nx

I I

MIXER

I.I.

AMPLIFIER

l

I

L

FIXED

DEMODULATOR

AMPLIFIER

AND
AUDIO

II

I I

1A4S5KC

VARIABLE

FIXED

11

OSCILLATOR

OSCILLATOR
I

CONVENTIONAL COMMUNICATIONS
RECEIVER

505 BC.

I

II HIGHLY SELECTIVE ACCESSORY
II AMPLIFIER AND DEMODULATOR

I

F.

(Q5'ER)I

_JL
Figure

10

TYPICAL DOUBLE -CONVERSION SUPERHETERODYNE RECEIVERS
Illustrated at (A) is the basic circuit of a commercial double- conversion superheterodyne receiver. At (B) is
illustrated the application of on accessory sharp i -f channel for obtaining improved selectivity from a conventional communications receiver through the use of the double-conversion principle.

than half the
taken alone.

Gm

of either of the two tubes

The Cascode r -f amplifier, developed at the MIT Radiation
Laboratory during World War II,
is a low noise circuit employing a grounded
cathode triode driving a grounded grid triode,
as shown in figure 9C. The stage gain of such
a circuit is about equal to that of a pentode
tube, while the noise figure remains at the low
level of a triode tube. Neutralization of the
first triode tube is usually unnecessary below
50 Mc. Above this frequency, a definite improvement in the noise figure may be obtained
Through the use of neutralization. The neutralizing coil, LN, should resonate at the operating frequency with the grid -plate capacity of
the first triode tube.
The 6B(27A and 6BZ7 tubes are designed for
use in cascode circuits, and may be used to
good advantage in the 144 Mc. and 220 Mc. amateur bands (figure 9D). For operation at higher
frequencies, the 6A)4 tube is recommended.
The Cascode

Amplifier

As previously mentioned,
the use of a higher intermediate frequency will also improve the image

Double Conversion

ratio, at the expense of i -f selectivity, by
placing the desired signal and the image farther apart. To give both good image ratio at
the higher frequencies and good selectivity in
the i -f amplifier, a system known as double
conversion is sometimes employed. In this system, the incoming signal is first converted to
a rather high intermediate frequency, and then
amplified and again converted, this time to a
much lower frequency. The first intermediate
frequency supplies the necessary wide separation between the image and the desired signal, while the second one supplies the bulk of
the i -f selectivity.
The double -conversion system, as illustrated in figure 10, is receiving two general
types of application at the present time. The
first application is for the purpose of attaining
extremely good stability in a communications
receiver through the use of crystal control of

the first oscillator. In such an arrangement,
as used in several types of Collins receivers,
the first oscillator is crystal controlled and is
followed by a tunable i -f amplifier which then
is followed by a mixer stage and a fixed-tuned
i -f a m p l i f i e r on a touch lower frequency.
Through such a circuit arrangement the sta-

bility of the complete receiver is equal to the

www.americanradiohistory.com

214

Radio

Receiver

THE

Fundamentals

RADIO

stability of the oscillator which feeds the second mixer, while the selectivity is determined
by the bandwidth of the second, fixed i -f am-

plifier.

The second common application of the
double- conversion principle is for the purpose
of obtaining a very high degree of selectivity
in the complete communications receiver. In
this type of application, as illustrated in figure 10 (B), a conventional communications receiver is modified in such a manner that its
normal i -f amplifier (which usually is in the
450 to 915 kc. range) instead of being fed to
a demodulator and then to the audio system,
is alternatively fed to a fixed -tune mixer stage
and then into a much lower intermediate frequency amplifier before the signal is demodulated and fed to the audio system. The accessory i -f amplifier system (sometimes called a
Q5'er) normally is operated on a frequency of
175 kc., 85 kc., or 50 kc.

12 -6

Signal- Frequency
Tuned Circuits

The signal- frequency tuned circuits in high frequency superheterodynes and tuned radio
frequency types of receivers consist of coils
of either the solenoid or universal -wound types
shunted by variable capacitors. It is in these
tuned circuits that the causes of success or
failure of a receiver often lie. The universal wound type coils usually are used at frequencies below 2000 kc.; above this frequency the
single-layer solenoid type of coil is more

satisfactory.
The two factors of greatest significance in determining the gain per -stage and selectivity, respectively, of a tuned amplifier are tuned- circuit
impedance and tuned -circuit Q. Since the resistance of modern capacitors is low at ordinary frequencies, the resistance usually can
be considered to be concentrated in the coil.
The resistance to be considered in making Q
determinations is the r -f resistance, not the
d -c resistance of the wire in the coil. The latter ordinarily is low enough that it may be
neglected. The increase in r -f resistance over
d -c resistance primarily is due to skin effect
and is influenced by such factors as wire size
and type, and the proximity of metallic objects
or poor insulators, such as coil forms with
high losses. Higher values of Q lead to better
selectivity and increased r -f voltage across
the tuned circuit. The increase in voltage is
due to an increase in the circuit impedance
with the higher values of Q.
Impedance

and Q

R F

C

INPUT

Figure 11
ILLUSTRATING "COMMON POINT"
BY- PASSING
To reduce the detrimental effects of cathode circuit inductance in v -h -f stages, all by -pass capacitors should be returned to the cathode terminal
at the socket. Tubes with two cathode leads can

give improved performance if the grid return is
made to one cathode terminal while the plate and
screen by -pass returns are made to the cathode
terminal which is connected to the suppressor
within the tube.

Frequently it is possible to secure an increase in impedance in a resonant circuit, and
consequently an increase in gain from an amplifier stage, by increasing the reactance
through the use of larger coils and smaller
tuning capacitors (higher L/C ratio).
Another factor which influences the operation of
tuned circuits is the input resistance of the
tubes placed across these circuits. At broadcast frequencies, the input resistance of most
conventional r -f amplifier tubes is high enough
so that it is not bothersome. But as the frequency is increased, the input resistance becomes lower and lower, until it ultimately
reaches a value so low that no amplification
can be obtained from the r -f stage.
The two contributing factors to the decrease
in input resistance with increasing frequency
are the transit time required by an electron
traveling between the cathode and grid, and
the inductance of the cathode lead common
to both the plate and grid circuits. As the
frequency becomes higher, the transit time
can become an appreciable portion of the time
required by an r -f cycle of the signal voltage,
and current will actually flow into the grid.
The result of this effect is similar to that
which would be obtained by placing a resistance between the tube's grid and cathode.
Input Resistance

Because the oscillator in a
superheterodyne operate s
"offset" from the other front
end circuits, it is necessary to make special
provisions to allow the oscillator to track

Superheterodyne

Tracking

when

similar tuning capacitor sections are

www.americanradiohistory.com

Tuning Circuits

HANDBOOK

215

MIXER
PADDING CAPACITOR

TUNING CAPACITOR

OSCILLATOR

SERIES TRACKING CAPACITOR

Figure 13
BANDSPREAD CIRCUITS
Parallel bandspread is illustrated at (A) and (B),
series bandspread at (C), and tq,ped.coil band-

Figure 12
SERIES TRACKING EMPLOYED
IN THE H -F OSCILLATOR OF A
SUPERHETERODYNE
The series tracking capacitor permits the use of
identical gangs in a ganged capacitor, since the
tracking capacitor slows down the rate of frequency change in the oscillator so that a constant difference in frequency between the oscillator and
the r -f stage (equal to the i -f amplifier frequency)
may be maintained.

ganged. The usual method of obtaining good
tracking is to operate the oscillator on the
high- frequency side of the mixer and use a
series tracking capacitor to slow down the
tuning rate of the oscillator. The oscillator
tuning rate must be slower because it covers
a smaller range than does the mixer when both
are expressed as a percentage of frequency.
At frequencies above 7000 kc. and with ordinary intermediate frequencies, the difference
in percentage between the two tuning ranges
is so small that it may be disregarded in receivers designed to cover only a small range,
such as an amateur band.
A mixer and oscillator tuning arrangement

in which a series tracking capacitor is provided is shown in figure 12. The value of the

tracking capacitor varies considerably with
different intermediate frequencies and tuning
ranges, capacitances as low as .0001 pfd.
being used at the lower tuning -range frequencies, and values up to .01 µfd. being used at
the higher frequencies.
Superheterodyne receivers designed to cover
only a single frequency range, such as the
standard broadcast band, sometimes obtain
tracking between the oscillator and the r -f circuits by cutting the variable plates of the oscillator tuning section to a different shape
from those used to tune the r -f stages.
frequency to which a
receiver responds may be
varied by changing the size
of either the coils or the capacitors in the tuning circuits, or both. In short -wave receivers
Frequency Range
Selection

The

spread at (D),

combination of both methods is usually employed, the coils being changed from one band
to another, and variable capacitors being used
to tune the receiver across each band. In practical receivers, coils may be changed by one
of two methods: a switch, controllable from
the panel, may be used to switch coils of different sizes into the tuning circuits or, alternatively, coils of different sizes may be
plugged manually into the receiver, the connection into the tuning circuits being made by
suitable plugs on the coils. Where there are
several plug -in cods for each band, they are
sometimes arranged to a single mounting strip,
allowing them all to be plugged in simultaneously.
a

In receivers using large tuning
capacitors to cover the shortwave spectrum with a minimum
of coils, tuning is likely to be quite difficult,
owing to the large frequency range covered by
a small rotation of the variable capacitors.
To alleviate this condition, some method of
slowing down the tuning rate, or bandspread ing, must be used.
Bandspread
Tuning

Quantitatively, bandspread is usually designated as being inversely proportional to the
range covered. Thus, a large amount of bandspread indicates that a small frequency range
is covered by the bandspread control. Conversely, a small amount of bandspread is taken
to mean that a large frequency range is covered
by the bandspread dial.
Types of
Bandspread

Bandspreading systems are of
two general types: electrical and

mechanical. Mechanical systems
are exemplified by high -ratio dials in which
the tuning capacitors rotate much more slowly

www.americanradiohistory.com

216

Radio

Receiver

THE

Fundamentals

than the dial knob. In this system, there is
often a separate scale or pointer either connected or geared to the dial knob to facilitate
accurate dial readings. However, there is a
practical limit to the amount of mechanical
bandspread which can be obtained in a dial
and capacitor before the speed- reduction unit
and capacitor bearings become prohibitively
expensive. Hence, most receivers employ a
combination of electrical and mechanical band spread. In such a system, a moderate reduction in the tuning rate is obtained in the dial,
and the rest of the reduction obtained by elec-

trical bandspreading.
In this book and in other radio
literature, mention is sometimes
made of stray or circuit capacitance. This capacitance is in the usual sense
defined as the capacitance remaining across
a coil when all the tuning, bandspread, and
padding capacitors across the circuit are at
their minimum capacitance setting.
Circuit capacitance can be attributed to two
general sources. One source is that due to the
input and output capacitance of the tube when
its cathode is heated. The input capacitance
varies somewhat from the static value when
the tube is in actual operation. Such factors
as plate load impedance, grid bias, and frequency will cause a change in input capacitance. However, in all except the extremely
high -transconductance tubes, the published
measured input capacitance is reasonably close
to the effective value when the tube is used
within its recommended frequency range. But
in the high -transconductance types the effective capacitance will vary considerably from
the published figures as operating conditions
are changed.
The second source of circuit capacitance,
and that which is more easily controllable, is
that contributed by the minimum capacitance
of the variable capacitors across the circuit
and that due to capacitance between the wiring and ground. In well -designed high -frequency receivers, every effort is made to keep
this portion of the circuit capacitance at a
minimum since a large capacitance reduces
the tuning range available with a given coil
and prevents a good L/C ratio, and consequently a high- impedance tuned circuit, from
being obtained.
A good percentage of stray circuit capacitance is due also to distributed capacitance
of the coil and capacitance between wiring
Stray Circuit

Capacitance

points and chassis.
Typical values of circuit capacitance may
run from 10 to 75 µpfd. in high- frequency re-

ceivers, the first figure representing concentric-line receivers with acorn or miniature
tubes and extremely small tuning capacitors,

RADIO

latter representing all -wave sets with
bandswitching, large tuning capacitors, and
conventional tubes.
and the

12 -7

I

-F Tuned Circuits

I -f amplifiers usually employ bandpass circuits of some sort. A bandpass circuit is exactly what the name implies -a circuit for passing a band of frequencies. Bandpass arrange-

ments can be designed for almost any degree
of selectivity, the type used in any particular
case depending upon the ultimate application
of the amplifier.
I.F
Transformers

Intermediate frequency trans formers ordinarily consist of
two or more tuned circuits and
some method of coupling the tuned circuits
together. Some representative arrangements
are shown in figure 14. The circuit shown at
A is the conventional i -f transformer, with the
coupling, M, between the tuned circuits being
provided by inductive coupling from one coil
to the other. As the coupling is increased, the
selectivity curve becomes less peaked, and
when a condition known as critical coupling
is reached, the top of the curve begins to flatten out. When the coupling is increased still
more, a dip occurs in the top of the curve.
The windings for this type of i -f transformer,
as well as most others, nearly always consist
of small, flat universal -wound pies mounted
either on a piece of dowel to provide an air
core or on powdered -iron for iron core i-f transformers. The iron -core transformers generally
have somewhat more gain and better selectivity
than equivalent air-core units.
The circuits shown at figure 14 -B and C are
quite similar. Their only difference is the type
of mutual coupling used, an inductance being
used at B and a capacitance at C. The operation of both circuits is similar. Three resonant circuits are formed by the components. In
B, for example, one resonant circuit is formed
by
C1, C, and L2 all in series. The frequency of this resonant circuit is just the same
as that of a single one of the coils and capacitors, since the coils and capacitors are similar in both sides of the circuit, and the resonant frequency of the two capacitors and the
two coils all in series is the same as that of
a single coil and capacitor. The second resonant frequency of the complete circuit is determined by the characteristics of each half of
the circuit containing the mutual coupling device. In B, this second frequency will be lower
than the first, since the resonant frequency of
C, and the inductance, M, or L2, C, and M
is lower than that of a single coil and capaci-

L

L

www.americanradiohistory.com

HANDBOOK

I

tor, due to the inductance of M being added to
the circuit.
The opposite effect takes place at figure
14 -C, where the common coupling impedance
is a capacitor. Thus, at C the second resonant frequency is higher than the first. In either
case, however, the circuit has two resonant frequencies, resulting in a flat- topped selectivity
curve. The width of the top of the curve is
controlled by the reactance of the mutual
coupling component. As this reactance is increased (inductance made greater, capacitance
made smaller), the two resonant frequencies
become further apart and the curve is broadened.
In the circuit of figure 14 -D, there is inductive coupling between the center coil and each
of the outer coils. The result of this arrangement is that the center coil acts as a sharply
tuned coupler between the other two. A signal
somewhat off the resonant frequency of the
transformer will not induce as much current
in the center coil as will a signal of the correct frequency. When a smaller current is induced in the center coil, it in turn transfers
a still smaller current to the output coil. The
effective coupling between the outer coils increases as the resonant frequency is approached, and remains nearly constant over a
small range and then decreases again as the
resonant band is passed.
Another very satisfactory bandpass arrangement, which gives a very straight- sided, flat topped curve, is the negative- mutual arrangement shown at figure 14 -E. Energy is transferred between the input and output circuits in
this arrangement by both the negative- mutual
coils, M, and the common capacitive reactance,
C. The negative- mutual coils are interwound
on the same form, and connected backward.
Transformers usually are made tunable over
a small range to permit accurate alignment in
the circuit in which they are employed. This
is accomplished either by means of a variable
capacitor across a fixed inductance, or by
means of a fixed capacitor across a variable
inductance. The former usually employ either
mica -compression capacitor (designated
a
"mica tuned "), or a small air dielectric variable capacitor (designated "air tuned"). Those
which use a fixed capacitor usually employ a
powdered iron core on a threaded rod to vary
the inductance, and are known as "permea-

bility tuned."
It is obvious that to pass modulation sidebands and to allow
for slight drifting of the transmitter carrier frequency and the receiver local oscillator, the
i -f amplifier must pass not a single frequency
but a band of frequencies. The width of this
pass band, usually 5 to 8 kc. at maximum

Shape Factor

-F

M

Amplifiers

217

M

E
Figure 14
-F AMPLIFIER COUPLING
ARRANGEMENTS
The interstoge coupling arrangements illustrated
above give a better shape factor (more straight
sided selectivity curve) than would the some number of tuned circuits coupled by means of tubes.
I

width in

a good communications receiver, is
known as the pass band, and is arbitrarily
taken as the width between the two frequencies at which the response is attenuated 6 db,
or is "6 db down." However, it is apparent
that to discriminate against an interfering signal which is stronger than the desired signal,
much more than 6 db attenuation is required.
The attenuation arbitrarily taken to indicate
adequate discrimination against an interfering

signal is 60 db.

www.americanradiohistory.com

218

Radio

Receiver

THE

Fundamentals

L

R

C

Figure

RADIO

16

ELECTRICAL EQUIVALENT OF
QUARTZ FILTER CRYSTAL
The crystal is equivalent to o very large value

of

inductance in series with small values of capacitance and resistance, with a larger though still
small value of capacitance across the whole circuit Yrepresenting holder capacitance plus stray
capacitances).

Figure 15
-F PASS BAND OF TYPICAL
COMMUNICATIONS RECEIVER
I

It is apparent that it is desirable to have
bandwidth at 60 db down as narrow as
possible, but it must be done without making
the pass band (6 db points) too narrow for satisfactory reception of the desired signal. The
figure of merit used to show the ratio of bandwidth at 6 db down to that at 60 db down is
designated shape factor. The ideal i -f curve,
a rectangle, would have a shape factor of 1.0.
The i -f shape factor in typical communications
receivers runs from 3.0 to 5.5.
The most practicable method of obtaining a
low shape factor for a given number of tuned
circuits is to employ them in pairs, as in figure 14 -A, adjusted to critical coupling (the
value at which two resonance points just begin to become apparent). If this gives too
sharp a "nose" or pass band, then coils of
lower Q should be employed, with the coupling
maintained at the critical value. As the Q is
lowered, closer coupling will be required for
critical coupling.
Conversely if the pass band is too broad,
coils of higher Q should be employed, the
coupling being maintained at critical. If the
pass band is made more narrow by using looser
coupling instead of raising the Q and main taninig critical coupling, the shape factor will
not be as good.
The pass band will not be much narrower
for several pairs of identical, critically coupled
tuned circuits than for a single pair. However,
the shape factor will be greatly improved as
each additional pair is added, up to about 5
pairs, beyond which the improvement for each
additional pair is not significant. Commercially available communications receivers of
the

good quality normally employ

3 or 4 double
tuned transformers with coupling adjusted to
critical or slightly less.
The pass band of a typical communication
receiver having a 455 kc. i -f amplifier is shown
in figure 15.

"Miller
Effect"

As mentioned previously, the dynamic input capacitance of a tube varies
slightly with bias. As a -v-c voltage
normally is applied to i -f tubes for radiotelephony reception, the effective grid-cathode
capacitance varies as the signal strength
varies, which produces the same effect as
slight detuning of the i -f transformer. This
effect is known as "Miller effect," and can
be minimized to the extent that it is not
troublesome either by using a fairly low L/C
ratio in the transformers or by incorporating
a small amount of degenerative feedback, the
latter being most easily accomplished by leaving part of the cathode resistor unbypassed
for r.f.
Crystal Filters

The pass band of an intermediate frequency amplifier
may be made very narrow through the use of a
piezoelectric filter crystal employed as a
series resonant circuit in a bridge arrangement known as a crystal filter. The shape factor is quite poor, as would be expected when
the selectivity is obtained from the equivalent
of a single tuned circuit, but the very narrow
pass band obtainable as a result of the extremely high Q of the crystal makes the crystal filter useful for c -w telegraphy reception.
The pass band of a 455 kc. crystal filter may
be made as narrow as 50 cycles, while the
narrowest pass band that can be obtained with
a 455 kc. tuned circuit of practicable dimensions is about 5 kc.
The electrical equivalent of a filter crystal
is shown in figure 16. For a given frequency,
L is very high, C very low, and R (assuming

www.americanradiohistory.com

Filters

Crystal

HANDBOOK

219

CRYSTAL

E

SELECTIVITY
CONTROL

PHASING
CONTROL

Figure

For

a

Figure 17
EQUIVALENT OF CRYSTAL
FILTER CIRCUIT
given voltage out of the generator, the volt-

age developed across Z1 depends upon the ratio
of the impedance of X to the sum of the impedances

of Z and Z1. Because of the high

of the crystal,
its impedance changes rapidly with changes in
Q

frequency.

a good crystal of high Q) is very low. Capacitance C, represents the shunt capacitance of

the electrodes, plus the crystal holder and
wiring, and is many times the capacitance of
C. This makes the crystal act as a parallel
resonant circuit with a frequency only slightly
higher than that of its frequency of series
resonance. For crystal filter use it is the
series resonant characteristic that we are primarily interested in.
The electrical equivalent of the basic crystal filter circuit is shown in figure 17. If the
impedance of Z plus Z, is low compared to the
impedance of the crystal X at resonance, then
the current flowing through
and the voltage
developed across it, will be almost in inverse
proportion to the impedance of X, which has
a very sharp resonance curve.
If the impedance of Z plus Z, is made high
compared to the resonant impedance of X, then
there will be no appreciable drop in voltage
across Z, as the frequency departs from the
resonant frequency of X until the point is
reached where the impedance of X approaches
that of Z plus Z,. This has the effect of broadening out the curve of frequency versus voltage
developed across
which is another way of
saying that the selectivity of the crystal filter
(but not the crystal proper) has been reduced.
In practicable filter circuits the impedances
Z and Z, usually are represented by some form
of tuned circuit, but the basic principle of
operation is the same.

Z

Z

Fractical Filters

It is necessary to balance

out the capacitance across
the crystal holder (C in figure 16) to prevent
bypassing around the crystal undesired signals
off the crystal resonant frequency. The balancing is done by a phasing circuit which
takes out -of-phase voltage from a balanced in-

18

TYPICAL CRYSTAL FILTER CIRCUIT

put circuit and passes it to the output side of
the crystal in proper phase to neutralize that
passed through the holder capacitance. A rep-

resentative practical filter arrangement is
shown in figure 18. The balanced input circuit
may be obtained either through the use of a
split- stator capacitor as shown, or by the use
of a center -tapped input coil.
Variable- Selec-

circuit of figure 18, the
selectivity is minimum with
the crystal input circuit tuned
to resonance, since at resonance the impedance of the tuned circuit is maximum. As the
input circuit is detuned from resonance, however, the impedance decreases, and the selectivity becomes greater. In this circuit, the output from the crystal filter is tapped down on
the i -f stage grid winding to provide a low
value of series impedance in the output circuit. It will be recalled that for maximum selectivity, the total impedance in series with the
crystal (both input and output circuits) must
be low. If one is made low and the other is
made variable, then the selectivity may be
tivity Filters

In the

varied at will from sharp to broad.
The circuit shown in figure 19 also achieves
variable selectivity by adding a variable impedance in series with the crystal circuit. In
this case, the variable impedance is in series
with the crystal output circuit. The impedance
of the output circuit is varied by varying the
Q. As the Q is reduced (by adding resistance
in series with the coil), the impedance decreases and the selectivity becomes greater.
The input circuit impedance is made low by
using a non -resonant secondary on the input
transformer.
A variation of the circuit shown at figure
19 consists of placing the variable resistance
across the coil and capacitor, rather than in
series with them. The result of adding the resistor is a reduction of the output impedance,
and an increase in selectivity. The circuit behaves oppositely to that of figure 19, however;
as the resistance is lowered the selectivity
becomes greater. Still another variation of figure 19 is to use the tuning capacitor across
the output coil to vary the output impedance.

www.americanradiohistory.com

220

Radio

Receiver Fundamentals

RADIO

THE

CRYSTwL

SELECTIVITY
CONTROL

Figure

r

CRYSTAL NOTCH

+I

+2 +3

z
3 3S

19

VARIABLE SELECTIVITY
CRYSTAL FILTER

O

This circuit permits of a greater control of selectivity than does the circuit of figure 16, and does
not require a split- stator variable capacitor.

m

0 40
J

V
CI

-3

-I

-2

so
455

+

KC

Figure 20
-F PASS BAND OF TYPICAL
CRYSTAL FILTER
COMMUNICATIONS RECEIVER

As the output circuit is detuned from resonance, its impedance is lowered, and the
selectivity increases. Sometimes a set of
fixed capacitors and a multipoint switch are
used to give step -by -step variation of the output circuit tuning, and thus of the crystal

filter selectivity.
As previously discussed, a filter
crystal has both a resonant(series
resonant) and an anti -resonant
(parallel resonant) frequency, the impedance
of the crystal being quite low at the former
frequency, and quite high at the latter frequency. The anti- resonant frequency is just
slightly higher than the resonant frequency,
the difference depending upon the effective
shunt capacitance of the filter crystal and
holder. As adjustment of the phasing capacitor
controls the effective shunt capacitance of the
crystal, it is possible to vary the anti -resonant frequency of the crystal slightly without
unbalancing the circuit sufficiently to let undesired signals leak through the shunt capacitance in appreciable amplitude. At the exact
anti -resonant frequency of the crystal the attenuation is exceedingly high, because of the
high impedance of the crystal at this frequency. This is called the rejection notch, and
can be utilized virtually to eliminate the
heterodyne image or repeat tuning of c -w signals. The beat frequency oscillator can be
so adjusted and the phasing capacitor so adjusted that the desired beat note is of such
a pitch that the image (the same audio note
on the other side of zero beat) falls in the rejection notch and is inaudible. The receiver
then is said to be adjusted for single -signal
Rejection
Notch

operation.
The rejection notch sometimes can be employed to reduce interference from an undesired phone signal which is very close in
frequency to a desired phone signal. The filter
is adjusted to "broad" so as to permit tele-

I

phony reception, and the receiver tuned so
that the carrier frequency of the undesired
signal falls in the rejection notch. The modulation sidebands of the undesired signal still
will come through, but the carrier heterodyne

will

be effectively eliminated and interference
greatly reduced.
A typical crystal selectivity curve for a
communications receiver is shown in figure 20.

Crystal Filter
Considerations

A

crystal

filter, especially

when adjusted for single sig-

nal reception, greatly reduces
interference and background noise, the latter
feature permitting signals to be copied that
would ordinarily be too weak to be heard above
the background hiss. However, when the filter
is adjusted for maximum selectivity, the pass
band is so narrow that the received signal
must have a high order of stability in order to
stay within the pass band. Likewise, the local
oscillator in the receiver must be highly stable,
or constant retuning will be required. Another
effect that will be noticed with the filter adjusted too "sharp" is a tendency for code
characters to produce a ringing sound, and
have a hangover or "tails." This effect limits
the code speed that can be copied satisfactorily when the filter is adjusted for extreme

selectivity.
The Collins Mechanical Fil ter (figure 21) is a new concept in the field of selectivity. It is an electro- mechanical bandpass
filter about half the size of a cigarette package. As shown in figure 22, it consists of an
input transducer, a resonant mechanical secThe Mechanical

Filter

www.americanradiohistory.com

Collins Mechanical Filter

HANDBOOK
tion comprised of a number of metal discs, and
an output transducer.
The frequency characteristics of the resonant mechanical section provide the almost
rectangular selectivity curves shown in figure
23. The input and output transducers serve
only as electrical to mechanical coupling devices and do not affect the selectivity characteristics which are determined by the metal
discs. An electrical signal applied to the input terminals is converted into a mechanical
vibration at the input transducer by means of
magnetostriction. This mechanical vibration
travels through the resonant mechanical section to the output transducer, where it is converted by magnetostriction to an electrical
signal which appears at the output terminals.
In order to provide the most efficient electromechanical coupling, a small magnet in the
mounting above each transducer applies a magnetic bias to the nickel transducer core. The
electrical impulses then add to or subtract
from this magnetic bias, causing vibration of
the filter elements that corresponds to the
exciting signal. There is no mechanical motion
except for the imperceptible vibration of the
metal discs.
Magnetostrictively -driven mechanical filters
have several advantages over electrical equivalents. In the region from 100 kc. to 500 kc.,
the mechanical elements are extremely small,
and a mechanical filter having better selectivity than the best of conventional i -f systems
may be enclosed in a package smaller than one
i -f transformer.
Since mechanical elements with Q's of 5000
or more are readily obtainable, mechanical filters may be designed in accordance with the
theory for lossless elements. This permits filter characteristics that are unobtainable with
electrical circuits because of the relatively
high losses in electrical elements as compared
with the mechanical elements used in the

221

c.

Figure 21
COLLINS MECHANICAL FILTERS
The
Collins Mechanical Filter is an
electro- mechanical bandpass filter which
surpasses, in one small unit, the selectivity of conventional, space-consuming
filters. At the left is the miniaturized
filter, less than 2!4' long. Type H is
next, and two horizontal mounting types
are at right. For exploded view of Collins
Mechanical Filter, see figure 46.

The frequency characteristics of the mechanical filter are permanent, and no adjustment is
required or is possible. The filter is enclosed
in a hermetically sealed case.
In order to realize full benefit from the mechanical filter's selectivity characteristics,
it is necessary to provide shielding between
the external input and output circuits, capable

of reducing transfer of energy external to the

o

filters.

ONE SUPPORTING
DISC AT

EACH END

RESONANT

MECHANICAL SECTION

(0 RESONANT DISCS)

ill

COUPLING RODS
DIAS MAGNET

iì%U
MAGNETOSTRICTIVE

DRIVING ROD

RANSDUCER

COIL

ELECTRICAL SIGNAL
(INPUT OR OUTPUT)

ELECTRICAL SIGNAL
(INPUT OR OUTPUT)

Figure 22
MECHANICAL FILTER

FUNCTIONAL DIAGRAM

Figure 23
Selectivity curves of 4554c. mechanical filters
with nominal 0.8 -%c. (dotted line) and 3.1 -kc.
(solid line) bandwidth at -6 db.

www.americanradiohistory.com

222

Radio Receiver

THE

Fundamentals
I VERY SMALL

65J7

I.F. STAGE

RADIO

DET.
AUDIO

//1

Por

I-rlp 2Np
I

DETECTOR
BY

J
pA

I.F STAGE

Figure 24
A

GRID LEAK DETECTOR
DET.

VARIABLE -OUTPUT B -F -O CIRCUIT
beat- frequency oscillator whose output is con-

trollable is of considerable assistance in copying
c-w signals over a wide ronge of levels, and such
a control is almost a necessity for satisfactory
copying of single -sideband radiophone signals.
AUDIO

filter by a minimum value of 100 db. If the input circuit is allowed to couple energy into
the output circuit external to the filter, the
excellent skirt selectivity will deteriorate and
the passband characteristics will be distorted.
As with almost any mechanically resonant
circuit, elements of the mechanical filter have
multiple resonances. These result in spurious
modes of transmission through the filter and
produce minor passbands at frequencies on
other sides of the primary passband. Design
of the filter reduces these sub -bands to a low
level and removes them from the immediate
area of the major passband. Two conventional
i -f transformers supply increased attenuation
to these spurious responses, and are sufficient
to reduce them to an insignificant level.
The beat -frequency oscillator,
usually called the b.J.o., is a
necessary adjunct for reception of c -w telegraph signals on superheterodynes which have no other provision for obtaining modulation of an incoming c -w telegraphy signal. The oscillator is coupled into
or just ahead of second detector circuit and
supplies a signal of nearly the same frequency
as that of the desired signal from the i -f amplifier. If the i -f amplifier is tuned to 455 kc.,
for example, the b.f.o. is tuned to approximately 454 or 456 kc. to produce an audible
(1000 cycle) beat note in the output of the
second detector of the receiver. The carrier
signal itself is, of course, inaudible. The b.f.o.
is not used for voice reception, except as an
aid in searching for weak stations.
The b -f -o input to the second detector need
only be sufficient to give a good beat note on
an average signal. Too much coupling into the
second detector will give an excessively high
hiss level, masking weak signals by the high
noise background.
Figure 24 shows a method of manually ad-

OB

DIODE DETECTOR

I.F. STAGE

DET.

AUDIO

©

PLATE DETECTOR

I.F. STAGE

DET.

Beat- Frequency

Oscillators

OD

INFINITE IMPEDANCE DETECTOR

Figure 25

TYPICAL CIRCUITS FOR GRID -LEAK,
DIODE, PLATE AND INFINITE IMPEDANCE DETECTOR STAGES

justing the b -f -o output to correspond with the
strength of received signals. This type of variable b -f -o output control is a useful adjunct
to any superheterodyne, since it allows sufficient b-f-o output to be obtained to beat with
strong signals or to allow single - sideband
reception and at the same time permits the
b -f -o output, and consequently the hiss, to be
reduced when attempting to receive weak signals. The circuit shown is somewhat better
than those in which one of the electrode volt-

www.americanradiohistory.com

HANDBOOK

Detector Circuits

TYPICAL

A -V -C

223

Figure 26
CIRCUIT USING

A DOUBLE DIODE
Any of the small dual-diode tubes may be used in
this circuit. Or, if desired, a duo- diodetriode may be used, with the triode acting as the first audio
stage. The left-hand diode serves
as the detector, while the right -hand side
acts as the a-v -c rectifier. The use of separate
diodes for detector and o-v -c reduces distortion when
receiving an AM signal with a high

modulation percentage.

b -f-o tube is changed, as the latter
circuits usually change the frequency of the

ages on the

b.f.o. at the same time they change the strength,
making it necessary to reset the trimmer each
time the output is adjusted.
The b.f.o. usually is provided with a small
trimmer which is adjustable from the front
panel to permit adjustment over a range of 5
or 10 kc. For single -signal reception the b.f.o.
always is adjusted to the high- frequency side,
in order to permit placing the heterodyne image
in the rejection notch.
In order to reduce the b -f -o signal output
voltage to a reasonable level which will prevent blocking the second detector, the signal
voltage is delivered through a low- capacitance
(high -reactance) capacitor having a value of
1 to 2 fgifd.
Care must be taken with the b.f.o. to prevent harmonics of the oscillator from being
picked up at multiples of the b -f-o frequency.
The complete b.f.o. together with the coupling
circuits to the second detector, should be thoroughly shielded to prevent pickup of the harmonics by the input end of the receiver.
If b-f-o harmonics still have a tendency to
give trouble after complete shielding and isolation of the b -f-o circuit has been accomplished, the passage of these harmonics from
the b-f-o circuit to the rest of the receiver can
be stopped through the use of a low -pass filter
in the lead between the output of the b -f -o circuit and the point on the receiver where the
b-f-o signal is to be injected.
12 -8

Detector, Audio, and
Control Circuits

Detectors

Second detectors for use in superheterodynes are usually of the

diode, plate, or infinite-impedance types. Occasionally, grid-leak detectors are used in receivers using one i-f stage or none at all, in
which case the second detector usually is
made regenerative.
Diodes are the most popular second detectors because they allow a simple method of
obtaining automatic volume control to be used.
Diodes load the tuned circuit to which they
are connected, however, and thus reduce the
selectivity slightly. Special i -f transformers
are used for the purpose of providing a low impedance input circuit to the diode detector.
Typical circuits for grid -leak, diode, plate
and infinite -impedance detectors are shown
in figure 25.
The elements of an automatic
volume control (a.v.c.) system are shown in figure 26.
A dual -diode tube is used as a combination
diode detector and a -v -c rectifier. The left hand diode operates as a simple rectifier in
the manner described earlier in this chapter.
Audio voltage, superimposed on a d -c voltage,
appears across the 500,000 -ohm potentiometer
(the volume control) and the .0001 -µfd. capacitor, and is passed on to the audio amplifier.
The right -hand diode receives signal voltage
directly from the primary of the last i -f amplifier, and acts as the a -v -c rectifier. The pulsating d-c voltage across the 1- megohm a.v.c.diode load resistor is filtered by a 500,000 -ohm
resistor and a .05-pfd. capacitor, and applied
as bias to the grids of the r -f and i -f amplifier
tubes; an increase or decrease in signal
strength will cause a corresponding increase
or decrease in a-v -c bias voltage, and thus the
gain of the receiver is automatically adjusted
to compensate for changes in signal strength.
Automatic Volurne Control

www.americanradiohistory.com

224

Radio

A -C Loading of
Second Detector

Receiver

THE

Fundamentals

By disassociating the a.v.c.
and detecting functions

through using separate
diodes, as shown, most of the ill effects of
a-c shunt loading on the detector diode are
avoided. This type of loading causes serious
distortion, and the additional components required to eliminate it are well worth their cost.
Even with the circuit shown, a -c loading can
occur unless a very high (5 megohms, or more)
value of grid resistor is used in the following
audio amplifier stage.
A.V.C. in

IF

IF

RF

RADIO

RFoR IF

In receivers having

a beat frequency oscillator for the
reception of radiotelegraph
signals, the use of a.v.c.
can result in a great loss in sensitivity when
the b.f.o. is switched on. This is because the
beat oscillator output acts exactly like a
strong received signal, and causes the a -v -c
circuit to put high bias on the r -f and i -f stages,
thus greatly reducing the receiver's sensitivity. Due to the above effect, it is necessary to
provide a method of making the a -v -c circuit
inoperative when the b.f.o. is being used. The
simplest method of eliminating the a -v -c action is to short the a -v-c line to ground when
the b.f.o. is turned on. A two -circuit switch
may be used for the dual purpose of turning on
the beat oscillator and shorting out the a.v.c.
if desired.

B-F-O- Equipped
Receivers

Visual means for determining
whether or not the receiver is
properly tuned, as well as an
indication of the relative signal strength, are
both provided by means of tuning indicators
(S meters) of the meter or vacuum -tube type.
A d -c milliammeter can be connected in the
plate supply circuit of one or more r -f or i -f
amplifiers, as shown in figure 27A, so that the
change in plate current, due to the action of
the a -v -c voltage, will be indicated on the instrument. The d -c instrument MA should have
a full -scale reading approximately equal to the
total plate current taken by the stage or stages
whose plate current passes through the instrument. The value of this current can be estimated by assuming a plate current on each
stage (with no signal input to the receiver) of
Signal Strength

R

Fon

I

F

+70

6U5/6G5
OR

6E5

TO A V C

Indicators

about 6 ma. However, it will be found to be
more satisfactory to measure the actual plate
current on the stages with a milliammeter of
perhaps 0 -100 ma. full scale before purchasing
an instrument for use as an S meter. The
50 -ohm potentiometer shown in the drawing is
used to adjust the meter reading to full scale
with no signal input to the receiver.
When an ordinary meter is used in the plate
circuit of a stage, for the purpose of indicating signal strength, the meter reads backwards

t250v

Figure 27
SIGNAL -STRENGTH -METER CIRCUITS
Shown above are four circuits for obtaining a signal- strength reading which is a function of incoming carrier amplitude. The circuits are discussed
lin the accompanying text.

with respect to strength. This is because increased a -v -c bias on stronger signals causes
lower plate current through the meter. For this
reason, special meters which indicate zero at
the right -hand end of the scale are often used
for signal strength indicators in commercial
receivers using this type of circuit. Alternatively, the meter may be mounted upside
down, so that the needle moves toward the
right with increased strength.
The circuit of figure 27B can frequently be
used to advantage in a receiver where the cathode of one of the r-f or i -f amplifier stages
runs directly to ground through the cathode
bias resistor instead of running through a cath-

www.americanradiohistory.com

HANDBOOK
ode -voltage gain control. In this case a 0 -1
d -c milliammeter in conjunction with a resistor
from 1000 to 3000 ohms can be used as shown
as a signal- strength meter. With this circuit
the meter will read backwards with increasing
signal strength as in the circuit previously

discussed.

Figure 27C is the circuit of a forward-readingS meter as is often used in communications
receivers. The instrument is used in an unbalanced bridge circuit with the d -c plate resistance of one i -f tube as one leg of the
bridge and with resistors for the other three
legs. The value of the resistor R must be determined by trial and error and will be somewhere in the vicinity of 50,000 ohms. Sometimes the screen circuits of the r -f and i -f
stages are taken from this point along with
the screen-circuit voltage divider.
Electron -ray tubes (sometimes called "magic
eyes") can also be used as indicators of relative signal strength in a circuit similar to that
shown in figure 27D. A 6U5/6G5 tube should
be used where the a-v -c voltage will be from
5 to 20 volts and a type 6E5 tube should be
used when the a-v-c voltage will run from 2
to 8 volts.

amplifiers are employed in nearly all radio
receivers. The audio amplifier stage or stages
are usually of the Class A type, although Class
AB push -pull stages are used in some receivers. The purpose of the audio amplifier is
to bring the relatively weak signal from the
detector up to a strength sufficient to operate
a pair of headphones or a loud speaker. Either
triodes, pentodes, or beam tetrodes may be
used, the pentodes and beam tetrodes usually
giving greater output. In some receivers, particularly those employing grid leak detection,
it is possible to operate the headphones directly from the detector, without audio amplification. In such receivers, a single audio
stage with a beam tetrode or pentode tube is
ordinarily used to drive the loud speaker.
Most communications receivers, either home constructed or factory -made, have a single ended beam tetrode (such as a 6L6 or 6V6) or
pentode (6F6 or 6K6 -GT) in the audio output
stage feeding the loudspeaker. If precautions
are not taken such a stage will actually bring
about a decrease in the effective signal -tonoise ratio of the receiver due to the rising
high- frequency characteristic of such a stage
when feeding a loud -speaker. One way of improving this condition is to place a mica or
paper capacitor of approximately 0.003 pfd.
capacitance across the primary of the output
transformer. The use of a capacitor in this
manner tends to make the load impedance seen
by the plate of the output tube more constant
Audio Ampl if iers

Audio

Noise

Suppression

225

over the audio -frequency range. The speaker
and transformer will tend to present a rising
impedance to the tube as the frequency increases, and the parallel capacitor will tend
to make the total impedance more constant
since it will tend to present a decreasing impedance with increasing audio frequency.
A still better way of improving the frequency
characteristic of the output stage, and at the
same time reducing the harmonic distortion,
is to use shunt feedback from the plate of the
output tube to the plate of a tube such as a
6SJ7 acting as an audio amplifier stage ahead
of the output stage.

Noise Suppression

12-9

The problem of noise suppression confronts
the listener who is located in places where
interference from power lines, electrical appliances, and automobile ignition systems is
troublesome. This noise is often of such intensity as to swamp out signals from desired

stations.
There are two principal methods for reducing this noise:
(1) A-c line filters at the source of interference, if the noise is created by an
electrical appliance.
(2) Noise -limiting circuits for the reduction, in the receiver itself, of interference of the type caused by automobile
ignition systems.

appliances, such
as electric mixers, heating pads,
vacuum sweepers, refrigerators,
oil burners, sewing machines, doorbells, etc.,
create an interference of an intermittent nature. The insertion of a line filter near the
source of interference often will effect a complete cure. Filters for small appliances can
consist of a 0.1 -µfd. capacitor connected across the 110 -volt a -c line. Two capacitors
in series across the line, with the midpoint
connected to ground, can be used in conjunction with ultraviolet ray machines, refrigerators, oil burner furnaces, and other more stubborn offenders. In severe cases of interference, additional filters in the form of heavy duty r -f choke coils must be connected in
series with the 110 -volt a -c line on both sides
of the line right at the interfering appliance.
Power Line

Filters

Many household

Numerous noise -limiting circuits
which are beneficial in overcoming key clicks, automobile ignition interference, and similar noise impulses
have become popular. They operate on the
Peak Noise

Limiters

principle that each individual noise pulse is

226

Radio

of very short duration, yet of very high amplitude. The popping or clicking type of noise
from electrical ignition systems may produce
a signal having a peak value ten to twenty
times as great as the incoming radio signal,
but an average power much less than the signal.
As the duration of this type of noise peak
is short, the receiver can be made inoperative
during the noise pulse without the human ear
detecting the total loss of signal. Some noise
limiters actually punch a bole in the signal,
while others merely limit the maximum peak
signal which reaches the headphones or loud-

speaker.
The noise peak is of such short duration
that it would not be objectionable except for
the fact that it produces an over -loading effect
on the receiver, which increases its time constant. A sharp voltage peak will give a kick
to the diaphragm of the headphones or speaker, and the momentum or inertia keeps the
diaphragm in motion until the dampening of
the diaphragm stops it. This movement produces a popping sound which may completely
obliterate the desired signal. If the noise pulse
can be limited to a peak amplitude equal to
that of the desired signal, the resulting interference is practically negligible for moderately low repetition rates, such as ignition
noise.
In addition, the i -f amplifier of the receiver
will also tend to lengthen the duration of the
noise pulses because the relatively high -Q i -f
tuned circuits will ring or oscillate when excited by a sharp pulse, such as produced by
ignition noise. The most effective noise limiter
would be placed before the high -Q i -f tuned
circuits. At this point the noise pulse is the
sharpest and has not been degraded by passage through the i -f transformers. In addition,
the pulse is eliminated before it can produce
ringing effects in the i -f chain.
noise limiter is shown in
figure 28. This is an adaptation of the Lamb noise silencer circuit. The i-f signal is fed into a double
grid tube, such as a 6L7, and thence into the
i -f chain. A 6AB7 high gain pentode is capacity coupled to the input of the i -f system.
This auxiliary tube amplifies both signal and
noise that is fed to it. It has a minimum of
selectivity ahead of it so that it receives the
true noise pulse before it is degraded by the
i -f strip. A broadly tuned i -f transformer is used
to couple the noise amplifier to a 6H6 noise
rectifier. The gain of the noise amplifier is
controlled by a potentiometer in the cathode
of the 6AB7 noise amplifier. This potentiometer controls the gain of the noise amplifier

The Lomb
Noise Limiter

THE

Receiver Fundamentals

An i -f

IST

DET

RADIO

ISTI.F.

2ND

I.F.

617

Figure 28
THE LAMB I -F NOISE SILENCER

stage and in addition sets the bias level on
the 6H6 diode so that the incoming signal will
not be rectified. Only noise peaks louder than
the signal can overcome the resting bias of
the 6H6 and cause it to conduct. A noise pulse
rectified by the 6H6 is applied as a negative
voltage to the control grid of the 6L7 i -f tube,
disabling the tube, and punching a hole in the
signal at the instant of the noise pulse. By
varying the bias control of the noise limiter,
the negative control voltage applied to the
6L7 may be adjusted until it is barely sufficient to overcome the noise impulses applied
to the al control grid without allowing the
modulation peaks of the carrier to become
badly distorted.

effective i -f noise
limiter is the Bishop limiter.
This is a full -wave shunt type
diode limiter applied to the primary of the last
i -f transformer of a receiver. The limiter is
self- biased and automatically adjusts itself
to the degree of modulation of the received
signal. The schematic of this limiter is shown
in figure 29. The bias circuit time constant is
determined by C, and the shunt resistance,
The Bishop

Another

Noise Limiter

which consists of R, and R2 in series. The
plate resistance of the last i -f tube and the
capacity of C, determine the charging rate of
the circuit. The limiter is disabled by opening
which allows the bias to rise to the value
of the i -f signal.

S

www.americanradiohistory.com

HANDBOOK

Noise Limiters

227

ter peak noise suppression than a standard
communications receiver having an i -f bandwidth of perhaps 8 kc. Likewise, when a crystal filter is used on the "sharp" position an
a -f peak' limiter is of little benefit.

Practical

Noise limiters range all the
way from an audio stage
Limiter Circuits
running at very low screen
or plate voltage, to elaborate affairs employing 5 or more tubes. Rather
than attempt to show the numerous types, many
of which are quite complex considering the
results obtained, only two very similar types
will be described. Either is just about as effective as the most elaborate limiter that can
be constructed, yet requires the addition of
but a single diode and a few resistors and
capacitors over what would be employed in a
good superheterodyne without a limiter. Both
circuits, with but minor modifications in resistance and capacitance values, are incorporated in one form or another in different
types of factory -built communications receivPeak Noise

Figure 29
THE BISHOP I -F NOISE LIMITER

Audio Noise

Limiters

of the simplest and most
practical peak limiters for radioSome

telephone reception employ one
or two diodes either as shunt or series limiters
in the audio system of the receiver. when a
noise pulse exceeds a certain predetermined
threshold value, the limiter diode acts either
as a short or open circuit, depending upon
whether it is used in a shunt or series circuit.
The threshold is made to occur at a level high
enough that it will not clip modulation peaks
enough to impair voice intelligibility, but low
enough to limit the noise peaks effectively.
Because the action of the peak limiter is
needed most on very weak signals, and these
usually are not strong enough to produce proper
a -v-c action, a threshold setting that is correct for a strong phone signal is not correct
for optimum limiting on very weak signals. For
this reason the threshold control often is tied
in with the a -v -c system so as to make the
optimum threshold adjustment automatic instead of manual.
Suppression of impulse noise by means of
an audio peak limiter is best accomplished at
the very front end of the audio system, and
for this reason the function of superheterodyne
second detector and limiter often are combined
in a composite circuit.
The amount of limiting that can be obtained
is a function of the audio distortion that can
be tolerated. Because excessive distortion
will reduce the intelligibility as much as will
background noise, the degree of limiting for
which the circuit is designed has to be a compromise.
Peak noise limiters working at the second
detector are much more effective when the i -f
bandwidth of the receiver is broad, because a
sharp i -f amplifier will lengthen the pulses by
the time they reach the second detector, making the limiter less effective. V-h -f super heterodynes have an i -f bandwidth considerably wider than the minimum necessary for
voice sidebands (to take care of drift and instability). Therefore, they are capable of bet-

ers.

Referring to figure 30, the first circuit shows
conventional superheterodyne second detector, a.v.c., and first audio stage with the
addition of one tube element,
which may
be either a separate diode or part of a twindiode as illustrated. Diode D, acts as a series
gate, allowing audio to get to the grid of the
a -f tube only so long as the diode is conducting. The diode is biased by a d -c voltage obtained in the same manner as a -v -c control voltage, the bias being such that pulses of short
duration no longer conduct when the pulse
voltage exceeds the carrier by approximately
60 per cent. This also clips voice modulation
peaks, but not enough to impair intelligibility.
It is apparent that the series diode clips
only positive modulation peaks, by limiting
upward modulation to about 60 per cent. Negative or downward peaks are limited automatically to 100 per cent in the detector, because
obviously the rectified voltage out of the
diode detector cannot be less than zero. Limiting the downward peaks to 60 per cent or so
instead of 100 per cent would result in but
little improvement in noise reduction, and the
results do not justify the additional components required.
It is important that the exact resistance
values shown be used, for best results, and
that 10 per cent tolerance resistors be used
for R, and R,. Also, the rectified carrier voltage developed across C, should be at least 5
volts for good limiting.
The limiter will work well on c -w telegraphy
if the amplitude of beat frequency oscillator
injection is not too high. Variable injection is
to be preferred, adjustable from the front panel.
a

D

228

THE

Receiver Fundamentals

Radio

V2

VI
LAST I. F.TUDC

I

-0.1 -µfd.

paper
mice
C_100 -44fd. mica
C4, Cs -0.01 -µfd. paper
megohm,
Rr, R2
15 watt
R,, R4- 220,000 ohms,
Vt watt
R6, R6-1 megohm,
C1

AUDIO

FT.

RADIO

C,

-50 -4µtd.

-1

4T

Rr

watt

-2- megohm potentiometer

Figure 30
NOISE LIMITER CIRCUIT, WITH ASSOCIATED A -V -C
This limiter is of the series type, and is self-adjusting to carrier strength for phone reception. For proper
operation several volts should be developed across the secondary of the last i -f transformer (IFT) under
carrier conditions.

If this feature is not provided, the b -f-o injection should be reduced to the lowest value that
will give a satisfactory beat. When this is
done, effective limiting and a good beat can
be obtained by proper adjustment of the r -f
and a -f gain controls. It is assumed, of course,
that the a.v.c. is cut out of the circuit for
c -w telegraphy reception.
Alternative
Limiter Circuit

The
more

circuit of figure 31 is
effective than that

shown in figure 30 under certain conditions and requires the addition of

only one more resistor and one more capacitor than the other circuit. Also, this circuit
involves a smaller loss in output level than
the circuit of figure 30. This circuit can be
used with equal effectiveness with a combined diode - triode or diode -pentode tube (6R7,
6SR7, 6Q7, 6SQ7 or similar diode -triodes, or
6B8, 6SF7, or similar diode -pentodes) as diode
detector and first audio stage. However, a
separate diode must be used for the noise
limiter, D,. This diode may be one -half of a
6H6, 6AL5, 7A6, etc., or it may be a triode
connected 6J5, 6C4 or similar type.
Note that the return for the volume control
must be made to the cathode of the detector
diode (and not to ground) when a dual tube is
used as combined second- detector first- audio.
This means that in the circuit shown in figure
31 a connection will exist across the points
where the "X" is shown on the diagram since
a common cathode lead is brought out of the
tube for Dr and V,. If desired, of course, a
single dual diode may be used for Dr and D,
in this circuit as well as in the circuit of figure 30. Switching the limiter in and out with
the switch S brings about no change in volume.
In any diode limiter circuit such as the ones
shown in these two figures it is important that

the mid -point of the heater potential for the
noise -limiter diode be as close to ground
potential as possible. This means that the
center -tap of the heater supply for the tubes
should be grounded wherever possible rather
than grounding one side of the heater supply
as is often done. Difficulty with hum pickup
in the limiter circuit may be encountered when
one side of the heater is grounded due to the
high values of resistance necessary in the

limiter circuit.
The circuit of figure

31 has been used with
excellent success in several home -constructed
receivers, and in the BC- 312/BC -342 and BC348 series of surplus communications receivers. It is also used in certain manufactured
receivers.
An excellent check on the operation of the
noise limiter in any communications receiver
can be obtained by listening to the Loran signals in the 160 -meter band. With the limiter
out a sharp rasping buzz will be obtained
when one of these stations is tuned in. With
the noise limiter switched into the circuit the
buzz should be greatly reduced and a low pitched hum should be heard.

The most satisafctory diode
noise limiter is the series full wave limiter, shown in figure
32. The positive noise peaks are clipped by
diode A, the clipping level of which may be
adjusted to clip at any modulation level between 25 per cent and 100 per cent. The negaative noise peaks are clipped by diode B at

The Full -Wave

Limiter

a

fixed level.

Twin Noise Squelch.
popularized by CQ magazine, is a combination of a diode noise clipper
and an audio squelch tube. The squelch cit.-

The TNS

Limiter

The

HANDBOOK

U

-H -F

Circuits

229

This circuit is of the selfadjusting type and gives less
distortion for a given degree
of modulation than the more

limiter circuits.
-470K, 14 won
R3 -100K,
watt

common
R1, R2

R.,

RS

V2

-1

megohm, VI watt
megohm potentiometer

-2mica (approx.)
c2- 0.01 -µtd. paper
R6

C1- 0.00025

C3- 0.01 -4fd.
C4- 0.01 -12fd.

paper
paper

02 -6116,

D3,

6AL5,
sections of

diode

7A6,
a

or
6S8 -GT

Figure

31

ALTERNATIVE NOISE LIMITER CIRCUIT

cuit is useful in eliminating the grinding background noise that is the residual left by the
diode clipper. In figure 33, the setting of the
470K potentiometer determines the operating
level of the squelch action and should be set
to eliminate the residual background noise.
Because of the low inherent distortion of the
TNS, it may be left in the circuit at all times.
As with other limiters, the TNS requires a
high signal level at the second detector for
maximum limiting effect.
12 -10

Special Considerations

wavelength sections of parallel conductors or
concentric transmission line are not only more
efficient but also become of practical dimensions.

Tuning
Short Lines

Tubes and tuning capacitors con netted to the open end of a transmission line provide a capacitance that makes the resonant length less than
a quarter wave -length. The amount of shortening for a specified capacitive reactance is
determined by the surge impedance of the line

in U -H -F Receiver Design

increasingly higher frequencies, it becomes progressively
more difficult to obtain a satisfactory amount of selectivity and impedance
from an ordinary coil and capacitor used as a

Transmission

At

F

STAGE

2ND DEI. -AUDIO

Line Circuits

S n. DIO

resonant circuit. On the other hand, quarter
F

T

2ND DET

AuD10

Figure 32
THE FULL -WAVE SERIES AUDIO
NOISE LIMITER

Figure 33
THE TNS AUDIO NOISE LIMITER

www.americanradiohistory.com

230

Radio

Receiver

Fundamentals

THE

RADIO

CAVITY

CAVITY

0LOOV
LINE

CONCENTRIC

LINE

CAVITY

Figure 34
COUPLING AN ANTENNA TO A

O

©

GVITY

CAVITY

GRIDS

ELECTRON
BEAM

HOLE

COAXIALRESONANT CIRCUIT
(A) shows the recommended method for coupling
a coaxial line to a coaxial resonant circuit. (B)
shows an alternative method for use with an open wire type of antenna feed line.

section. It is given by the equation for resonance:
1

2rr /C

=

Z. tan

1

77 = 3.1416, / is the frequency, C the
capacitance, Z. the surge impedance of the
line, and tan / is the tangent of the electrical
length in degrees.
The capacitive reactance of the capacitance
across the end is 1 /(277 / C) ohms. For resonance, this must equal the surge impedance of
the line times the tangent of its electrical
length (in degrees, where 90° equals a quarter wave). It will be seen that twice the capacitance will resonate a line if its surge impedance is halved; also that a given capacitance
has twice the loading effect when the frequency is doubled.

in which

It is possible to couple into
a parallel -rod line by tapping directly on one or both
rods, preferably through
blocking capacitors if any d.c. is present.
More commonly, however, a hairpin is inductively coupled at the shorting bar end, either
to the bar or to the two rods, or both. This
normally will result in a balanced load. Should
a loop unbalanced to ground be coupled in,
any resulting unbalance reflected into the rods
can be reduced with a simple Faraday screen,
made of a few parallel wires placed between
the hairpin loop and the rods. These should
be soldered at only one end and grounded.
An unbalanced tap on a coaxial resonant

Coupling Into
Lines and
Coaxial Circuits

Figure 35
METHODS OF EXCITING A RESONANT

CAVITY

circuit can be made directly on the inner conductor at the point where it is properly matched
(figure 34). For low impedances, such as a concentric line feeder, a small one -half turn loop
can be inserted through a hole in the outer
conductor of the coaxial circuit, being in effect a half of the hairpin type recommended
for coupling balanced feeders to coaxial resonant lines. The size of the loop and closeness to the inner conductor determines the
impedance matching and loading. Such loops
coupled in near the shorting disc do not alter
the tuning appreciably, if not overcoupled.
cavity is a closed resonant
made of metal. It is
known also as a rhumbatron. The
cavity, having both inductance and capacitance, supersedes coil -capacitor and capacitance- loaded transmission -line tuned circuits
at extremely high frequencies where conventional L and C components, of even the most
refined design, prove impractical because of
the tiny electrical and physical dimensions
they must have. Microwave cavities have high
Q factors and are superior to conventional
tuned circuits. They may be employed in the
manner of an absorption wavemeter or as the
tuned circuit in other r -f test instruments, and
in microwave transmitters and receivers.
Resonant cavities usually are closed on all
sides and all of their walls are made of electrical conductor. However, in some forms,
small openings are present for the purpose of
excitation. Cavities have been produced in
several shapes including the plain sphere,

Resonant

Cavities

A

chamber

www.americanradiohistory.com

HANDBOOK

TUNING

U

dimpled sphere, sphere with reentrant cones
of various sorts, cylinder, prism (including
cube), ellipsoid, ellipsoid -hyperboloid, doughnut- shape, and various reentrant types. In appearance, they resemble in their simpler forms
metal boxes or cans.
The cavity actually is a linear circuit, but
one which is superior to a conventional coaxial resonator in the s -h -f range. The cavity
resonates in much the same manner as does a
barrel or a closed room with reflecting walls.
Because electromagnetic energy, and the
associated electrostatic energy, oscillates to
and fro inside them in one mode or another,
resonant cavities resemble wave guides. The
mode of operation in a cavity is affected by
the manner in which micro-wave energy is injected. A cavity will resonate to a large number of frequencies, each being associated with
a particular mode or standing -wave pattern.
The lowest mode (lowest frequency of operation) of a cavity resonator normally is the one
used.
The resonant frequency of a cavity may be
varied, if desired, by means of movable plungers or plugs, as shown in figure 36A, or a
movable metal disc (see figure 36B). A cavity
that is too small for a given wavelength will
not oscillate.
The resonant frequencies of simple spherical, cylindrical, and cubical cavities may be
calculated simply for one particular mode.
Wavelength and cavity dimensions (in centimeters) are related by the following simple
resonance formulae:

Butterfly

231

DISC

Figure 36
TUNING METHODS FOR CYLINDRICAL
RESONANT CAVITIES

= 2.6 x
= 2.83 x
= 2.28x

radius
half of
radius

1

side

Unlike the cavity resonator, which
in its conventional form is a device
which can tune over a relatively
narrow band, the butterfly circuit is a tunable
resonator which permits coverage of a fairly
Circuit

Circuits

MOVEABLE

SLUGS

For Cylinder A,
" Cube
Ar
" Sphere A,

-H -F

Figure 37
THE BUTTERFLY RESONANT CIRCUIT
Shown at (A) is the physical appearance of the
butterfly circuit as used in the v -h -f and lower
u-h-f ronge. (B) shows an electrical representation of the circuit.

wide

u -h -f band. The butterfly circuit is very
similar to a conventional coil -variable capacitor combination, except that both inductance
and capacitance are provided by what appears
to be a variable capacitor alone. The Q of this
device is somewhat less than that of a concentric -line tuned circuit but is entirely adequate for numerous applications.
Figure 37A shows construction of a single
butterfly section. The butterfly- shaped rotor,
from which the device derives its name, turns
in relation to the unconventional stator. The
two groups of stator "fins" or sectors are in
effect joined together by a semi -circular metal
band, integral with the sectors, which provides
the circuit inductance. When the rotor is set
to fill the loop opening (the position in which
it is shown in figure 37A), the circuit inductance and capacitance are reduced to minimum.
When the rotor occupies the position indicated
by the dotted lines, the inductance and capacitance are at maximum. The tuning range of
practical butterfly circuits is in the ratio of
1.5:1 to 3.5:1.
Direct circuit connections may be made to
points A and B. If balanced operation is desired, either point C or D will provide the
electrical mid-point. Coupling may be effected
by means of a small singleturn loop placed
near point E or F. The butterfly thus permits
continuous variation of both capacitance and
inductance, as indicated by the equivalent
circuit in figure 37B, while at the same time
eliminating all pigtails and wiping contacts.
Several butterfly sections may be stacked
in parallel in the same way that variable capacitors are built up. In stacking these sections, the effect of adding inductances in parallel is to lower the total circuit inductance,
while the addition of stators and rotors raises
the total capacitance, as well as the ratio of
maximum to minimum capacitance.

www.americanradiohistory.com

232

Radio Receiver Fundamentals

Butterfly circuits have been applied specifically to oscillators for transmitters, superheterodyne receivers, and heterodyne frequency meters in the 100 - 1000 -Mc. frequency range.
The types of resonant circuits described in the previous paragraphs
have largely replaced conventional
coil -capacitor circuits in the range above 100
Mc. Tuned short lines and butterfly circuits
are used in the range from about 100 Mc. to
perhaps 3500 Mc., and above about 3500 Mc.
resonant cavities are used almost exclusively.
The resonant cavity is also quite generally
employed in the 2000 -Mc. to 3500 -Mc. range.
In a properly designed receiver, thermal
agitation in the first tuned circuit is amplified
by subsequent tubes and predominates in the
output. For good signal-to- set -noise ratio,
therefore, one must strive for a high -gain low noise r -f stage. Hiss can be held down by giving careful attention to this point. A mixer
has about 0.3 of the gain of an r -f tube of the
same type; so it is advisable to precede a
mixer by an efficient r-f stage. It is also of
some value to have good r -f selectivity before
the first detector in order to reduce noises
produced by beating noise at one frequency
against noise at another, to produce noise at
the intermediate frequency in a superheterodyne.
The frequency limit of a tube is reached
when the shortest possible external connections are used as the tuned circuit, except for
abnormal types of oscillation. Wires or sizeable components are often best considered as
sections of transmission lines rather than as
simple resistances, capacitances, or inductReceiver

Circuits

ances.
as small triodes and pentodes will
operate normally, they are generally preferred
as v-h -f tubes over other receiving methods
that have been devised. However, the input
capacitance, input conductance, and transit
time of these tubes limit the upper frequency
at which they may be operated. The input resistance, which drops to a low value at very
short wave -lengths, limits the stage gain and
broadens the tuning.
So long

The first tube in a v -h -f receiver is
most important in raising the signal
above the noise generated in successive stages, for which reason small v -h-f types
are definitely preferred.
Tubes employing the conventional grid-controlled and diode rectifier principles have been
modernized, through various expedients, for
operation at frequencies as high, in some new
types, as 4000 Mc. Beyond that frequency,
electron transit time becomes the limiting facV -H -F

Tubes

THE

RADIO

tor and new principles must be enlisted. In
general, the improvements embodied in existing tubes have consisted of (1) reducing electrode spacing to cut down electron transit
time, (2) reducing electrode areas to decrease
interelectrode capacitances, and (3) shortening of electrode leads either by mounting the
electrode assembly close to the tube base or
by bringing the leads out directly through the
glass envelope at nearby points. Through reduction of lead inductance and interelectrode
capacitances, input and output resonant frequencies due to tube construction have bee':

increased substantially.
Tubes embracing one or more of the features just outlined include the later !octal
types, high- frequency acorns, button -base
types, and the lighthouse types. Type 6J4
button-base triode will reach 500 Mc. Type
6F4 acorn triode is recommended for use up to
1200 Mc. Type 1A3 button -base diode has a
resonant frequency of 1000 Mc., while type
9005 acorn diode resonates at 1500 Mc. Lighthouse type 2C40 can be used at frequencies
up to 3500 Mc. as an oscillator.
More than two de c a d e s have
passed since the crystal (mineral)
rectifier enjoyed widespread use
in radio receivers. Low -priced tubes completely supplanted the fragile and relatively insensitive crystal detector, although it did continue for a few years as a simple meter rectifier in absorption wavemeters after its demise
as a receiver component.
Today, the crystal detector is of new importance in microwave communication. It is
being employed as a detector and as a mixer
in receivers and test instruments used at ex-

Crystal

Rectifiers

tremely high radio frequencies. At some of
the frequencies employed in microwave operations, the crystal rectifier is the only satisfactory detector or mixer. The chief advantages of the crystal rectifier are very low capacitance, relative freedom from transit -time
difficulties, and its two -terminal nature. No
batteries or a -c power supply are required for
its operation.
The crystal detector consists essentially of
a small piece of silicon or germanium mounted in a base of low- melting -point alloy and
contacted by means of a thin, springy feeler
wire known as the cat whisker. This arrangement is shown in figure 38A.
The complex physics of crystal rectification
is beyond the scope of this discussion. It is
sufficient to state that current flows from several hundred to several thousand times more
readily in one direction through the contact of
cat whisker and crystal than in the opposite
direction. Consequently, an alternating current
(including one of microwave frequency) will

www.americanradiohistory.com

HANDBOOK

SYMBOL

f

Receiver Adjustment

\'i

BRASS BASE CONNECTOR

-CERAMIC SLEEVE
BRASS CAP
BRASS CONNECTOR PIN

CRYSTAL DIODE

small silicon crystal is attached to
the base connector and o fine "cat whisker" wire is set to the most sensitive spot on the crystal. After adjustment the ceramic shell is filled with
compound to hold the contact wire in
position. Crystals of this type are used
to over 30,000 mc.
A

rectified by the crystal detector. The load,
through which the rectified currents flow, may
be connected in series or shunt with the crystal, although the former connection is most
generally employed.
The basic arrangement of a modern fixed
crystal detector developed during World War II
for microwave work, particularly radar, is
shown in figure 38B. Once the cat whisker of
this unit is set at the factory to the most sensitive spot on the surface of the silicon crystal and its pressure is adjusted, a filler compound is injected through the filling hole to
hold the cat whisker permanently in position.
be

Receiver Adjustment

simple regenerative receiver requires
adjustment other than that necessary
to insure correct tuning and smooth regeneration over some desired range. Receivers of
the tuned radio- frequency type and superheterodynes require precise alignment to obtain the
highest possible degree of selectivity and
A

little

sensitivity.
Good results can be obtained from a receiver only when it is properly aligned and adjusted. The most practical technique for mak-

ing these adjustments is given below.
A very small number

ments

Alignment procedure in a multistage t -r -f receiver is exactly the same as aligning a
single stage. If the detector is regenerative,
each preceding stage is successively aligned
while keeping the detector circuit tuned to the
test signal, the latter being a station signal
or one locally generated by a test oscillator
loosely coupled to the antenna lead. During
these adjustments, the r -f amplifier gain control is adjusted for maximum sensitivity, assuming that the r-f amplifier is stable and
does not oscillate. Often a sensitive receiver
can be roughly aligned by tuning for maximum
noise pickup.
Alignment

1N23 MICROWAVE -TYPE

Instruments

receiver output when tuning to a modulated
signal. If the signal is a steady tone, such as
from a test oscillator, the output meter will
indicate the value of the detected signal. In
this manner, alignment results may be visually
noted on the meter.
T -R -F Receiver

Figure 38

12 -11

233

of instru-

will suffice to check

and

align a communications receiver, the most important of these testing units being a modulated oscillator and a d -c and a -c voltmeter.
The meters are essential in checking the voltage applied at each circuit point from the power supply. If the a -c voltmeter is of the oxide rectifier type, it can be used, in addition, as
an output meter when connected across the

Aligning a superhet is a detailed task requiring a great
amount of care and patience.
It should never be undertaken without a thorough understanding of the involved job to be
done and then only when there is abundant

Superheterodyne
Alignment

time to devote to the operation. There are no
short cuts; every circuit must be adjusted individually and accurately if the receiver is to
give peak performance. The precision of each
adjustment is dependent upon the accuracy
with which the preceding one was made.
Superhet alignment requires (1) a good signal generator (modulated oscillator) covering
the radio and intermediate frequencies and
equipped with an attenuator; (2) the necessary
socket wrenches, screwdrivers, or "neutralizing tools" to adjust the various i -f and r -f
trimmer capacitors; and (3) some convenient
type of tuning indicator, such as a copper oxide or electronic voltmeter.
Throughout the alignment process, unless
specifically stated otherwise, the r -f gain control must be set for maximum output, the beat
oscillator switched off, and the a.v.c. turned
off or shorted out. When the signal output of
the receiver is excessive, either the attenuator
or the a -f gain control may be turned down, but
never the r -f gain control.

After the receiver has been
given a rigid electrical and
mechanical inspection, and any faults which
may have been found in wiring or the selection and assembly of parts corrected, the i -f
amplifier may be aligned as the first step in
the checking operations.
Vl ith the signal generator set to give a modulated signal on the frequency at which the i -f
I

-F Alignment

www.americanradiohistory.com

234

amplifier is to operate, clip the "hot" output
lead from the generator to the last i -f stage
through a small fixed capacitor to the control
grid. Adjust both trimmer capacitor's in the
last i -f transformer (the one between the last
i -f amplifier and the second detector) to resonance as indicated by maximum deflection of
the output meter.
Each i -f stage is adjusted in the same manner, moving the hot lead, stage by stage, back
toward the front end of the receiver and backing off the attenuator as the signal strength
increases in each new position. The last adjustment will be made to the first i -f transformer, with the hot signal generator lead connected to the control grid of the mixer. Occasionally it is necessary to disconnect the
mixer grid lead from the coil, grounding it
through a 1,000- or 5,000 -ohm resistor, and
coupling the signal generator through a small
capacitor to the grid.
When the last i -f adjustment has been completed, it is good practice to go back through
the i -f channel, re- peaking all of the transformers. It is imperative that this recheck be
made in sets which do not include a crystal
filter, and where the simple alignment of the
i -f amplifier to the generator is final.

I

RADIO

THE

Radio Receiver Fundamentals
-F SIGNAL IN

ALONE
t

w

`

F

PLUS

O

MULTIPLIER

f

W
f

55

CC

FREQUENCY

Figure

39

THE Q- MULTIPLIER
The loss resistance of a high -Q
neutralized by regeneration
is

circuit
in a

feedback amplifier. A highly
selective passband is produced which
circuit of the
is coupled to the i -f
receiver.
simple

-F with
Crystal Filter

There are several ways of align ing an i -f channel which contains a crystal -filter circuit.
However, the following method is one which
has been found to give satisfactory results in
every case: An unmodulated signal generator
capable of tuning to the frequency of the filter
crystal in the receiver is coupled to the grid of
the stage which precedes the crystal filter in
the receiver. Then, with the crystal filter
switched in, the signal generator is tuned
slowly to find the frequency where the crystal
peaks. The receiver "S" meter may be used
as the indicator, and the sound heard from the
loudspeaker will be of assistance in finding
the point. When the frequency at which the
crystal peaks has been found, all the i -f transformers in the receiver should be touched up
to peak at that frequency.
I

Adjusting the beat oscillator on a receiver that has
no front panel adjustment is relatively simple.
It is only necessary to tune the receiver to
resonance with any signal, as indicated by
the tuning indicator, and then turn on the b.f.o.
B -F -Q

Adjustment

and set its trimmer (or trimmers) to produce
the desired beat note. Setting the beat oscil-

lator in this way will result in the beat note
being stronger on one "side" of the signal
than on the other, which is what is desired
for c -w reception. The b.f.o. should not be set
to zero beat when the receiver is tuned to

resonance with the signal, as this will cause
an equally strong beat to be obtained on both
sides of resonance.

Alignment of the front end of a
receiver is a
relatively simple process, consisting of first getting the oscillator to cover
the desired frequency range and then of peaking the various r -f circuits for maximum gain.
However, if the frequency range covered by
the receiver is very wide a fair amount of cut
and try will be required to obtain satisfactory
tracking between the r-f circuits and the oscillator. Manufactured communications receivers
should always be tuned in accordance with
the instructions given in the maintenance manual for the receiver.
Front -End

Alignment

12 -12

home -constructed

Receiving Accessories

The selectivity of a receiver
may be increased by raising
the Q of the tuned circuits of the i -f strip. A
simple way to accomplish this is to add a controlled amount of positive feedback to a tuned
circuit, thus increasing its Q. This is done in
the 0- multiplier, whose basic circuit is shown
in figure 39. The circuit L -C1 -C2 is tuned to
The Q- Multiplier

www.americanradiohistory.com

Receiving Accessories 235

HANDßCOK
I.F SIGNAL IN

I

-F SIGNAL OJT

.005

6
_

L2

Lt
.005
NULL

O

12 10X7

TO PLATE TERMINAL
OF FIRST I-F TUBE TNRU
2. OF COAXIAL LINE

PEAR

MULTIPLIER "NULL"

SELECTIfIrY

2 12Ax7

CONTROLS

3

32 MEG

1.SK

10K

I

LI. GRAY6'URNE

455 KC

FREQUENCY

Figure

MULTIPLIER NULL CIRCUIT

1-

n

T

6.3 V.

(0.6-6.OAIN)

intermediate frequency, and the loss resistance of the circuit is neutralized by the
positive feedback circuit composed of C3 and
the vacuum tube. Too great a degree of positive
feedback will cause the circuit to break into

LI is required to tune out the
reactance of the coaxial line. It is adjusted for maximum signal response.
LI may be omitted if the Q- multiplier
is connected to the receiver with o
short length of wire, and the i -f transformer within the receiver is retuned.
Coil

the

oscillation.

At the resonant frequency, the impedance of
the tuned circuit is very high, and when shunted
across an i -f stage will have little effect upon
the signal. At frequencies removed from resonance, the impedance of the circuit is low, resulting in high attenuation of the i -f signal.
The resonant frequency of the Q- multiplier may
by varied by changing the value of one of the
components in the tuned circuit.
The Q- multiplier may also be used to "null"
a signal by employing negative feedback to
control the plate resistance of an auxiliary
amplifier stage as shown in figure 40. Since the
grid- cathode phase shift through the Q- multiplier
is zero, the plate resistance of a second tube
may be readily controlled by placing it across
the Q- multiplier. At resonance, the high negative feedback drops the plate resistance of
V2, shunting the i -f circuit. Off resonance, the
feedback is reduced and the plate resistance
of V2 rises, reducing the amount of signal attenuation in the i -f strip. A circuit combining
both the "peak" and "null" features is shown
in figure 41.
A version of the common
mixer or converter stage

41

SCHEMATIC OF A 455KC
0- MULTIPLIER

The addition of a second triode permits
the 0- Multiplier to be used for nulling
out an unwanted hetrodyne.

The Product Detector

V.

L22GRAY6URNE "LOOPSTICK" COIL

Figure 40
Q-

+200-300

B

V6 CNOKE

A.

may be used as a second detector in a receiver
in place of the usual diode detector. The diode
is an envelope detector (section 12 -1) and develops a d -c output voltage from a single r -f
signal, and audio "beats" from two or more
input signals. A product detector (figure 42)
requires that a local carrier voltage be present

in order to produce an audio output signal.

R

PRODUCT
DETECTOR

-F SIGNAL

LOCAL
OSCILLATOR

Figure 42
THE PRODUCT
DETECTOR

Audio output signal is
when
developed
only
local oscillator is on.

www.americanradiohistory.com

HAUDIO OUTPUT

Radio Receiver Fundamentals

236

Va

V,

I

-F SIGNAL

VS

12AU7

12AU7
I-F

0,
r+AU01O OUT
4711

SIC.

SO

SEAT OSC.

SIGNAL

Figure

Figure 43
PENTAGRID MIXER
USED AS PRODUCT
DETECTOR

PRODUCT DETECTOR
VI and V2 act as cathode followers, delivering sideband signal
and local oscillator signal to
grounded grid triode mixer (V3).

Such a detector is useful for single sideband
work, since the inter -modulation distortion is

extremely low.
A pentagrid product detector is shown in
figure 43. The incoming signal is applied to
grid 3 of the mixer tube, and the local oscillator
is injected on grid 1. Grid bias is adjusted for
operation over the linear portion of the tube
characteristic curve. When grid 1 injection
is removed, the audio output from an unmodulated signal applied to grid 3 should be reduced
approximately 30 to 40 db below normal detection level. When the frequency of the local
oscillator is synchronized with the incoming
carrier, amplitude modulated signals may be
received by exalted carrier reception, wherein
the local carrier substitutes for the transmitted
carrier of the a -m signal.
Three triodes may be used as a product
detector (figure 44). Triodes V1 and V2 act
as cathode followers, delivering the sideband
signal and the local oscillator signal to a
grounded grid triode (V3) which functions as
the mixer stage. A third version of the product
detector is illustrated in figure 45. A twin
triode tube is used. Section V1 functions as
a cathode follower amplifier. Section V2 is a

.

?tl

ì

II14611

i.h°1

..

IIIIII

44

TRIPLE -TRIODE

Figure 45
DOUBLE -TRIODE
PRODUCT DETECTOR

"plate"

detector, the cathode of which is
with the cathode follower amplifier.
The local oscillator signal is injected into the
grid circuit of tube V2.
common

Figure 46
EXPLODED VIEW OF COLLINS
MECHANICAL FILTER

.1. 'If#I!IIItIYIIiIIIIvIIIII'I+II
vIIl°1i
www.americanradiohistory.com
1

1

CHAPTER THIRTEEN

Generation of
Radio Frequency Energy

A radio communication or broadcast transmitter consists of a source of radio frequency
power, or carrier; a system for modulating the
carrier whereby voice or telegraph keying or
other modulation is superimposed upon it; and
an antenna system, including feed line, for
radiating the intelligence- carrying radio frequency power. The power supply employed to
convert primary power to the various voltages
required by the r -f and modulator portions of
the transmitter may also be considered part
of the transmitter.
Voice modulation usually is accomplished
by varying either the amplitude or the frequency of the radio frequency carrier in accordance
with the components of intelligence to be
transmitted.
Radiotelegraph modulation (keying) normally is accomplished either by interrupting,
shifting the frequency of, or superimposing an
audio tone on the radio -frequency carrier in
accordance with the dots and dashes to be
transmitted.
The complexity of the radio- frequency generating portion of the transmitter is dependent
upon the power, order of stability, and frequency desired. An oscillator feeding an antenna
directly is the simplest form of radio- frequency
generator. A modern high -frequency transmitter,
on the other hand, is a very complex generator.
Such an equipment usually comprises a very

stable crystal -controlled or self-controlled
oscillator to stabilize the output frequency, a
series of frequency multipliers, one or more
amplifier stages to increase the power up to
the level which is desired for feeding the antenna system, and a filter system for keeping
the harmonic energy generated in the transmitter from being fed to the antenna system.

13

-1

Controlled
Oscillators

Self-

In Chapter Four, it was explained that the
amplifying properties of a tube having three
or more elements give it the ability to generate an alternating current of a frequency determined by the components associated with
it. A vacuum tube operated in such a circuit
is called an oscillator, and its function is
essentially to convert direct current into radio frequency alternating current of a predetermined frequency.
Oscillators for controlling the frequency of
conventional radio transmitters can be divided
into two general classes: self-controlled and

crystal- controlled.
There are a great many types of self-controlled oscillators, each of which is best suited

237

www.americanradiohistory.com

Generation of

238

OA SHUNT -FED HARTLEY

R

-F

Energy

THE

OB SHUNT -FED COLPITTS

R

©

RADIO

TUNED PLATE TUNED GRID

R

L+

250

Li

250
GRID
COIL

OD TUNED -PLATE UNTUNED GRID

EO

ELECTRON COUPLED

HO

CLAPP ELECTRON COUPLED

FO

COLPITTS ELECTRON COUPLED

L

OG CLAPP

Figure

1

COMMON TYPES OF SELF -EXCITED OSCILLATORS
Fixed capacitor values are typical, but will vary somewhat with the application.
In the Clapp oscillator circuits (G) and (H), capacitors Cr and C2 should have
reactance of 50 to 100 ohms at the operating frequency of the oscillator. Tuning ofa
these two oscillators is accomplished by capacitor C. In the circuits of (E), (F), and
(H), tuning of the tank circuit in the plate of the oscillator tube will have relatively
small effect on the frequency of oscillation. The plate tank circuit also may, if desired, be tuned to a harmonic of the oscillation frequency, or a broadly resonant
circuit may be used in this circuit position.

to a

particular application. They can further

be subdivided into the

classifications

of: neg-

ative -grid oscillators, electron -orbit oscillators, negative -resistance oscillators, velocity
modulation oscillators, and magnetron oscillators.

negative -grid oscillator is
a vacuum-tube amplifier with a sufficient portion of the output energy coupled back into the
input circuit to sustain oscillation. The conNegative -Grid

A

Oscillators

essentially

trol grid is biased negatively with respect to
the cathode. Common types of negative -grid
oscillators are diagrammed in figure 1.

Illustrated in figure 1 (A) is the
oscillator circuit which finds the
most general application at the present time;
this circuit is commonly called the Hartley.
The operation of this oscillator will be described as an index to the operation of all
negative -grid oscillators; the only real differThe Hartley

www.americanradiohistory.com

Oscillators

HANDBOOK
ence between the various circuits is the manner in which energy for excitation is coupled
from the plate to the grid circuit.
When plate voltage is applied to the Hartley
oscillator shown at (A), the sudden flow of
plate current accompanying the application of
plate voltage will cause an electro- magnetic
field to be set up in the vicinity of the coil.
The building -up of this field will cause a potential drdp to appear from turn-to -turn along
the coil. Due to the inductive coupling between the portion of the coil in which the plate
current is flowing and the grid portion, a potential will be induced in the grid portion.
Since the cathode tap is between the grid
and plate ends of the coil, the induced grid
voltage acts in such a manner as to increase
further the plate current to the tube. This action will continue for a short period of time
determined by the inductance and capacitance
of the tuned circuit, until the flywheel effect
of the tuned circuit causes this action to come
to a maximum and then to reverse itself. The
plate current then decreases, the magnetic
field around the coil also decreasing, until a
minimum is reached, when the action starts
again in the original direction and at a greater

amplitude than before. The amplitude of these
oscillations, the frequency of which is determined by the coil -capacitor circuit, will increase in a very short period of time to a limit
determined by the plate voltage of the oscil-

lator tube.

(8) shows a version of
oscillator. It can
be seen that this is essentially the same circuit as the Hartley except that the ratio of a
The Colpitts

Figure

1

the Colpitts

pair of capacitances in series determines the
effective cathode tap, instead of actually using a tap on the tank coil. Also, the net capacitance of these two capacitors comprises
the tank capacitance of the tuned circuit. This
oscillator circuit is somewhat less susceptible to parasitic (spurious) oscillations than
the Hartley.
For best operation of the Hartley and Colpitts oscillators, the voltage from grid to cathode, determined by the tap on the coil or the
setting of the two capacitors, normally should
be from 1/3 to 1/5 that appearing between
plate and cathode.
The tuned -plate tuned-grid oscillator illustrated at (C) has
a tank circuit in both the plate and grid circuits. The feedback of energy from the plate
to the grid circuits is accomplished by the
The T.P.T.G.

plate -to-grid inter -electrode capacitance within the tube. The necessary phase reversal in
feedback voltage is provided by tuning the
grid tank capacitor to the low side of the de-

239

sired frequency and the plate capacitor to the
high side. A broadly resonant coil may be substituted for the grid tank to form the T.N. T.
oscillator shown at (D).
Electron -Coupled

In any of the

oscillator circuits just described it is
possible to take energy from
the oscillator circuit by coupling an external
load to the tank circuit. Since the tank circuit
determines the frequency of oscillation of the
tube, any variations in the conditions of the
external circuit will be coupled back into the
frequency determining portion of the oscillator.
These variations will result in frequency instability.
The frequency determining portion of an
oscillator may be coupled to the load circuit
only by an electron stream, as illustrated in
(E) and (F) of figure 1. When it is considered
that the screen of the tube acts as the plate
to the oscillator circuit, the plate merely actOscillators

ing as a coupler to the load, then the similarity between the cathode -grid- screen circuit
of these oscillators and the cathode -grid -plate
circuits of the corresponding prototype can be
seen.
The electron- coupled oscillator has good
stability with respect to load and voltage variation. Load variations have a relatively small
effect on the frequency, since the only coupling between the oscillating circuit and the
load is through the electron stream flowing
through the other elements to the plate. The
plate is electrostatically shielded from the
oscillating portion by the bypassed screen.
The stability of the e.c.o. with respect to
variations in supply voltages is explained as
follows: The frequency will shift in one direction with an increase in screen voltage, while
an increase in plate voltage will cause it to
shift in the other direction. By a proper proportioning of the resistors that comprise the
voltage divider supplying screen voltage, it is
possible to make the frequency of the oscillator substantially independent of supply voltage variations.
The Clapp

Oscillator

relatively new type of oscillator
circuit which is capable of giving
excellent frequency stability is
A

illustrated in figure 1G. Comparison between
the more standard circuits of figure IA through
IF and the Clapp oscillator circuits of figures
1G and 1H will immediately show one marked
difference: the tuned circuit which controls
the operating frequency in the Clapp oscillator
is series resonant, while in all the more standard oscillator circuits the frequency controlling circuit is parallel resonant. Also, the
capacitors C, and C, are relatively large in
terms of the usual values for a Colpitts oscil-

240

Generation of

R

-F

THE

Energy

lator. In fact, the value of capacitors C, and
C, will be in the vicinity of 0.001 µfd. to
0.0025 µfd. for an oscillator which is to be
operated in the 1.8 -Mc. band.
The Clapp oscillator operates in the following manner: at the resonant frequency of the
oscillator tuned circuit (L, C) the impedance
of this circuit is at minimum (since it operates in series resonance) and maximum current flows through it. Note however, that C,
and C, also are included within the current
path for the series resonant circuit, so that at
the frequency of resonance an appreciable
voltage drop appears across these capacitors.
The voltage drop appearing across C, is applied to the grid of the oscillator tube as excitation, while the amplified output of the
oscillator tube appears across C, as the driving power to keep the circuit in oscillation.
Capacitors C, and C, should be made as
large in value as possible, while still permitting the circuit to oscillate over the full tuning range of C. The larger these capacitors
are made, the smaller will be the coupling between the oscillating circuit and the tube, and
consequently the better will be oscillator stability with respect to tube variations. High Gm
tubes such as the 6AC7, 6ÁG7, and 6CB6 will
permit the use of larger values of capacitance
at C, and C, than will more conventional tubes
such as the 6SJ7, 6V6, and such types. In general it may be said that the reactance of capacitors C, and C, should be on the order of
40 to 120 ohms at the operating frequency of
the oscillator -with the lower values of reactance going with high -Gm tubes and the
higher values being necessary to permit oscillation with tubes having Gm in the range of
2000 micromhos such as the 6SJ7.
It will be found that the Clapp oscillator
will have a tendency to vary in power output
over the frequency range of tuning capacitor
C. The output will be greatest where C is at
its largest setting, and will tend to fall off
with C at minimum capacitance. In fact, if
capacitors C, and C, have too large a value
the circuit will stop oscillation near the minimum capacitance setting of C. Hence it will
be necessary to use a slightly smaller value
of capacitance at C, and C, (to provide an increase in the capacitive reactance at this
point), or else the frequency range of the oscillator must be restricted by paralleling a fixed
capacitor across C so that its effective capacitance at minimum setting will be increased to
a value which will sustain oscillation.
In the triode Clapp oscillator, such as shown
at figure 1G, output voltage for excitation of
an amplifier, doubler, or isolation stage normally is taken from the cathode of the oscillator tube by capacitive coupling to the grid
of the next tube. However, where greater iso-

RADIO

lation of succeeding stages from the oscillating circuit is desired, the electron- coupled
Clapp oscillator diagrammed in figure 1H may
be used. Output then may be taken from the
plate circuit of the tube by capacitive coupling
with either a tuned circuit, as shown, or with
an r -f choke or a broadly resonant circuit in
the plate return. Alternatively, energy may be
coupled from the output circuit L,-C, by link
coupling. The considerations with regard to
and the grid tuned circuit are the same
as for the triode oscillator arrangement of

CC

figure 1G.

Negative- resistance oscillators often are used when
unusually high frequency
stability is desired, as in a frequency meter.
The dynatron of a few years ago and the newer
transitron are examples of oscillator circuits
which make use of the negative resistance
characteristic between different elements in
some multi -grid tubes.
In the dynatron, the negative resistance is a
consequence of secondary emission of electrons from the plate of a tetrode tube. By a
proper proportioning of the electrode voltage,
an increase in screen voltage will cause a
decrease in screen current, since the increased
screen voltage will cause the screen to attract
a larger number of the secondary electrons
emitted by the plate. Since the net screen current flowing from the screen supply will be
decreased by an increase in screen voltage,
it is said that the screen circuit presents a
negative resistance.
If any type of tuned circuit, or even a resistance- capacitance circuit, is connected in
series with the screen, the arrangement will
oscillate -provided, of course, that the external
circuit impedance is greater than the negative
resistance. A negative resistance effect similar to the dynatron is obtained in the transitron
circuit, which uses a pentode with the suppressor coupled to the screen. The negative resistance in this case is obtained from a combination of secondary emission and inter -electrode coupling, and is considerably more stable
than that obtained from uncontrolled secondary
emission alone in the dynatron. A representative transitron oscillator circuit is shown in
figure 2.
The chief distinction between a conventional negative grid oscillator and a negative
resistance oscillator is that in the former the
tank circuit must act as a phase inverter in
order to permit the amplification of the tube
to act as a negative resistance, while in the
latter the tube acts as its own phase inverter.
Thus a negative resistance oscillator requires
only an untapped coil and a single capacitor
Negative Resistance Oscillators

www.americanradiohistory.com

HANDBOOK

Oscillators

6SK7
TWO -TERMINAL

241

Figure 2
OSCILLATOR CIRCUITS

Both circuits may be used for an audio
oscillator or for frequencies into the
v -h -f range simply by placing a tank circuit tuned to the proper frequency where
indicated on the drawing. Recommended
values for the components are given below for both oscillators.

O
6SN7

TRANSITRON OSCILLATOR

OR

6J6

TRANSITION OSCILLATOR
0.01 -µfd. mica for r.f. 10 -µfd. elect. for c.f.
C2- 0.00005 -µfd. mica for r.f. 0.1 -4fd. paper for
a.f.
C3- 0.003-µfd. mico for r.f. 03-4fd. paper for
a.f.
C4- 001 -µfd. mica for r.f. 8 -4fd. elect. for a.f.
R3-220K S5 -watt carbon
R2 1800 ohms 55-watt carbon
R3 -22K 2 -watt carbon
R4 -22K 2 -watt carbon

C,-

CATHODE -COUPLED OSCILLATOR
C3- 0.00005 -µfd. mica for r.f. 0.1 -µfd. paper
for audio

C2- 0.003 -µfd.

mica for

audio

O

R1-47K
CATHODE COUPLED OSCILLATOR

as the frequency determining tank circuit, and
is classed as a two terminal oscillator. In fact,
the time constant of an R/C circuit may be

used as the frequency determining element and
such an oscillator is rather widely used as a

tunable audio frequency oscillator.
The Franklin oscillator makes
use of two cascaded tubes to
obtain the negative- resistance
effect (figure 3). The tubes may be either a
pair of triodes, tetrodes, or pentodes, a dual

The Fronklin
Oscillator

triode, or a combination of a triode and a multi grid tube. The chief advantage of this oscillator circuit is that the frequency determining
tank only has two terminals, and one side of
the circuit is grounded.
The second tube acts as a phase inverter to
give an effect similar to that obtained with the
dynatron or transitron, except that the effective
transconductance is much higher. If the tuned
circuit is omitted or is replaced by a resistor,
the circuit becomes a relaxation oscillator or
a multivibrator.
The Clapp oscillator has proved
to be inherently the most stable
of all the oscillator circuits discussed above, since minimum coupling between the oscillator tube and its associated
tuned circuit is possible. However, this inOscillator
Stability

R2

-1K

rd.

8 -pfd.

lect.

for

-watt carbon
-watt carbon
S5

1

herently good stability is with respect to tube
variations; instability of the tuned circuit with
respect to vibration or temperature will of
course have as much effect on the frequency of
oscillation as with any other type of oscillator
circuit. Solid mechanical construction of the
components of the oscillating circuit, along
with a small negative -coefficient compensating
capacitor included as an element of the tuned
circuit, usually will afford an adequate degree
of oscillator stability.

Figure 3
THE FRANKLIN OSCILLATOR CIRCUIT
A separate phase inverter tube is used in
this oscillator to feed a portion of the output
back to the input in the proper phase to sustain oscillation. The values of Cr and C2
should be as small as will permit oscillations
to be sustained over the desired frequency
range.

242

Generation of

R

-F

used to control the frequency of a transmitter in
which there are stringent
limitations on frequency tolerance, several precautions are taken to ensure that a variable
frequency oscillator will stay on frequency.
The oscillator is fed from a voltage regulated
power supply, uses a well designed and temperature compensated tank circuit, is of rugged
mechanical construction to avoid the effects
of shock and vibration, is protected against
excessive changes in ambient room temperature, and is isolated from feedback or stray
coupling from other portions of the transmitter
by shielding, filtering of voltage supply leads,
and incorporation of one or more buffer- amplifier stages. In a high power transmitter a small
amount of stray coupling from the final amplifier to the oscillator can produce appreciable
degradation of the oscillator stability if both
are on the same frequency. Therefore, the oscillator usually is operated on a subharmonic
of the transmitter output frequency, with one
or more frequency multipliers between the oscillator and final amplifier.
V. F.O.

Transmit -

When

ter Controls

13 -2

Quartz Crystal

Oscillators
Quartz is a naturally occuring crystal having a structure such that when plates are cut

relationships to the crystallographic axes, these plates will show the
piezoelectric effect -the plates will be deformed in the influence of an electric field,
and, conversely, when such a plate is compressed or deformed in any way a potential
difference will appear upon its opposite sides.
The crystal has mechanical resonance, and
will vibrate at a very high frequency because
of its stiffness, the natural period of vibration
depending upon the dimensions, the method of
electrical excitation, and crystallographic
orientation. Because of the piezoelectric properties, it is possible to cut a quartz plate
which, when provided with suitable electrodes,
will have the characteristics of a series resonant circuit with a very high L/C ratio and
very high Q. The Q is several times as high
as can be obtained with an inductor -capacitor
combination in conventional physical sizes.
The equivalent electrical circuit is shown in
figure 4A, the resistance component simply
being an acknowledgment of the fact that the
Q, while high, does not have an infinite value.
The shunt capacitance of the electrodes and
associated wiring (crystal holder and socket,
plus circuit wiring) is represented by the dotted portion of figure 4B. In a high frequency
in certain definite

THE

Energy

CI

RADIO

L

(SMALL)

Lt
(LARGE)

4: C2
I

(sraAr

SHUNT)

RI
(SMALL)

-J

Figure

4

EQUIVALENT ELECTRICAL CIRCUIT OF
QUARTZ PLATE IN A HOLDER
At (A) is shown the equivalent series -resonant circuit of the crystal itself, at (B) is
shown how the shunt capacitance of the
holder electrodes and associated wiring affects the circuit to the combination circuit
of (C) which exhibits both series resonance
and parallel resonance (anti -resonance), the
separation in frequency between the two
modes being very small and determined by
the ratio of C1 to C,.

crystal this will be considerably greater than
the capacitance component of an equivalent
series L/C circuit, and unless the shunt capacitance is balanced out in a bridge circuit,
the crystal will exhibit both resonant (series
resonant) and anti- resonant (parallel resonant)
frequencies, the latter being slightly higher
than the series resonant frequency and ap-

proaching it as C, is increased.
The series resonance characteristic is employed in crystal filter circuits in receivers
and also in certain oscillator circuits wherein
the crystal is used as a selective feedback
element in such a manner that the phase of the
feedback is correct and the amplitude adequate only at or very close to the series resonant frequency of the crystal.
While quartz, tourmaline, Rochelle salts,
ADP, and EDT crystals all exhibit the piezoelectric effect, quartz is the material widely
employed for frequency control.
As the cutting and grinding of quartz plates
has progressed to a high state of development
and these plates may be purchased at prices
which discourage the cutting and grinding by
simple hand methods for one's own use, the
procedure will be only lightly touched upon
here.
The crystal blank is cut from the raw quartz
at a predetermined orientation with respect to
the optical and electrical axes, the orientation
determining the activity, temperature coefficient, thickness coefficient, and other characteristics. Various orientations or "cuts" having useful characteristics are illustrated in
figure 5.

www.americanradiohistory.com

Crystal Oscillators

HANDBOOK

neo

243

T[.neror[ co rmus..T
eD fllT[eS
fe[0U[CT A7.11

OSCILLATORS

MK..

LOW feCWCMCT
AT
ST

Figure

5

CT

34.

¡

.N`

T

CT.DT,[T,fT

SSiS'

IT -ST.

L

er

-

Ta

b

2[1.10
Se

URUC

`«e.
s.ocs

,

ORIENTATION OF THE

y

COMMON CRYSTAL CUTS
Tóir[eTUet
Xff:
ó
veD.tl
^p.¢
xo.o
27e0

a` OSCILLATORS

L

1

D

-' >
1

'

a

r

ZERO
o

nA

OXillTOe3

.r.D filTCeS

p.c....,
cDC

The crystal blank is then rough -ground almost to frequency, the frequency increasing
in inverse ratio to the oscillating dimension
(usually the thickness). It is then finished to
exact frequency either by careful lapping, by
etching, or plating. The latter process consists of finishing it to a frequency slightly
higher than that desired and then silver plating
the electrodes right on the crystal, the frequency decreasing as the deposit of silver is
increased. If the crystal is not etched, it must
be carefully scrubbed and "baked" several
times to stabilize it, or otherwise the frequency and activity of the crystal will change with
time. Irradiation by X -rays recently has been
used in crystal finishing.
Unplated crystals usually are mounted in
pressure holders, in which two electrodes are
held against the crystal faces under slight
pressure. Unplated crystals also are sometimes mounted in an air -gap holder, in which
there is a very small gap between the crystal
and one or both electrodes. By making this
gap variable, the frequency of the crystal may
be altered over narrow limits (about 0.3% for

certain types).

The temperature coefficient of frequency for
various crystal cuts of the " -T" rotated family is indicated in figure 5. These angles are
typical, but crystals of a certain cut will vary
slightly. By controlling the orientation and dimensioning, the turning point (point of zero
temperature coefficient) for a BT cut plate may
be made either lower or higher than the 75 degrees shown. Also, by careful control of axes
and dimensions, it is possible to get AT cut
crystals with a very flat temperature- frequency

characteristic.

S,

CCnTUT

.T

f

..[ee

, .
_

lTtes

rr

-

[.

The first quartz plates used were either Y
cut or X cut. The former had a very high temperature coefficient which was discontinuous,
causing the frequency to jump at certain critical temperatures. The X cut had a moderately
bad coefficient, but it was more continuous,
and by keeping the crystal in a temperature
controlled oven, a high order of stability could
be obtained. However, the X cut crystal was
considerably less active than the Y cut, especially in the case of poorly ground plates.
For frequencies between 500 kc. and about
6 Mc., the AT cut crystal now is the most
widely used. It is active, can be made free
from spurious responses, and has an excellent
temperature characteristic. However, above
about 6 Mc. it becomes quite thin, and a difficult production job. Between 6 Mc. and about
12 Mc., the BT cut plate is widely used. It
also works well between 500 kc. and 6 Mc.,
but the AT cut is more desirable when a high
order of stability is desired and no crystal
oven is employed.
For low frequency operation on the order of
100 kc., such as is required in a frequency
standard, the GT cut crystal is recommended,
though CT and DT cuts also are widely used
for applications between 50 and 500 kc. The
CT, DT, and GT cut plates are known as contour cuts, as these plates oscillate along the
long dimension of the plate or bar, and are
much smaller physically than would be the
case for a regular AT or BT cut crystal for
the same frequency.
Crystal Holders

Crystals normally are purchased ready mounted. The

244

Generation of

R

-F

best type mount is determined by the type crystal and its application, and usually an optimum mounting is furnished with the crystal.
However, certain features are desirable in all
holders. One of these is exclusion of moisture
and prevention of electrode oxidization. The
best means of accomplishing this is a metal
holder, hermetically sealed, with glass insulation and a metal -to -glass bond. However, such
holders are more expensive, and a ceramic or
phenolic holder with rubber gasket will serve
where requirements are not too exacting.
Temperature Control;
Crystal Ovens

Where the frequency tolerance requirements are
not too stringent and the

ambient temperature does not include extremes,
an AT -cut plate, or a BT -cut plate with optimum (mean temperature) turning point, will
often provide adequate stability without resorting to a temperature controlled oven. However, for broadcast stations and other applications where very close tolerances must be
maintained, a thermostatically controlled oven,
adjusted for a temperature slightly higher than
the highest ambient likely to be encountered,
must of necessity be employed.

vibrating string can
vibrate on its harmonics, a quartz crystal will

Harmonic Cut

Just as

Crystals

be made to

THE

Energy

RADIO
EXCITATION

6J5

ETC

EXCITATION

+5

loo -isov.

BASIC PIERCE" OSCILLATOR

HOT -CATHODE -PIERCE"

OSCILLATOR

Figure 6
THE PIERCE CRYSTAL OSCILLATOR
CIRCUIT
Shown at (A) is the basic Pierce crystal oscillator circuit. A capacitance of 10 to 75
µtd. normally will be required at C1 for
optimum operation. If a plate supply voltage
higher thon indicated is to be used, RFC'
may be replaced by a 22,000 -ohm 2-watt resistor. Shown at (B) is an alternative arrangement with the r -f ground moved to the
plate, and with the cathode floating. This
alternative circuit has the advantage that
the full r-f voltage developed across the
crystal may be used os excitation to the next
stage, since one side of the crystal is
grounded.

a

exhibit mechanical résonance (and therefore
electrical resonance) at harmonics of its funda-

mental frequency. When employed in the usual
holder, it is possible to excite the crystal
only on its odd harmonics (overtones).
By grinding the crystal especially for harmonic operation, it is possible to enhance its
operation as a harmonic resonator. BT and AT
cut crystals designed for optimum operation
on the 3d, 5th and even the 7th harmonic are
available. The 5th and 7th harmonic types,
especially the latter, require special holder
and oscillator circuit precautions for satisfactory operation, but the 3d harmonic type
needs little more consideration than a regular
fundamental type. A crystal ground for optimum
operation on a particular harmonic may or may
not be a good oscillator on a different harmonic or on the fundamental. One interesting
characteristic of a harmonic cut crystal is that
its harmonic frequency is not quite an exact
multiple of its fundamental, though the disparity is very small.
The harmonic frequency for which the crystal was designed is the working frequency. It
is not the fundamental since the crystal itself
actually oscillates on this working frequency
when it is functioning in the proper manner.
When a harmonic -cut crystal is employed, a
selective tuned circuit must be employed somewhere in the oscillator in order to discrimi-

nate against the fundamental frequency or undesired harmonics. Otherwise the crystal might
not always oscillate on the intended frequency.
For this reason the Pierce oscillator, later
described in this chapter, is not suitable for
use with harmonic -cut crystals, because the
only tuned element in this oscillator circuit
is the crystal itself.

For a given crystal operating as an anti -resonant tank in a given oscillator at fixed load impedance and plate and
screen voltages, the r -f current through the
crystal will increase as the shunt capacitance
C2 of figure 4 is increased, because this effectively increases the step -up ratio of C, to Cr.
the crystal
For a given shunt capacitance,
current for a given crystal is directly proportional to the r -f voltage across C,. This voltage may be measured by means of a vacuum
tube voltmeter having a low input capacitance,
and such a measurement is a more pertinent
one than a reading of r -f current by means of
a thermogalvanometer inserted in series with
one of the leads to the crystal holder.
The function of a crystal is to provide accurate frequency control, and unless it is used
in such a manner as to take advantage of its
inherent high stability, there is no point in
using a crystal oscillator. For this reason a

Crystal Current;
Heating and Fracture

www.americanradiohistory.com

C

Crystal Oscillators

HANDBOOK
crystal oscillator should not be run at high
plate input in an attempt to obtain considerable power directly out of the oscillator, as
such operation will cause the crystal to heat,
with resultant frequency drift and possible
fracture.

13 -3

Crystal Oscillator

Circuits
Considerable confusion exists as to nomenclature of crystal oscillator circuits, due to a
tendency to name a circuit after its discoverer.
Nearly all the basic crystal oscillator circuits
were either first used or else developed independently by G. W. Pierce, but he has not been
so credited in all the literature.
Use of the crystal oscillator in master oscillator circuits in radio transmitters dates
back to about 1924 when the first application
articles appeared.
The Pierce The circuit of figure6A is the simplest crystal oscillator circuit. It
Oscillator
is one of those developed by

Pierce, and is generally known among amateurs as the Pierce oscillator. The crystal
simply replaces the tank circuit in a Colpitts
or ultra-audion oscillator. The r -f excitation
voltage available to the next stage is low, being somewhat less than that developed across
the crystal. Capacitor C, will make more of
the voltage across the crystal available for
excitation, and sometimes will be found necessary to ensure oscillation. Its value is small,
usually approximately equal to or slightly
greater than the stray capacitance from the
plate circuit to ground (including the grid of
the stage being driven).
If the r -f choke has adequate inductance, a
crystal (even a harmonic cut crystal) will almost invariably oscillate on its fundamental.
The Pierce oscillator therefore cannot be used
with harmonic cut crystals.
The circuit at (B) is the same as that of
(A) except that the plate instead of the cathode is operated at ground r -f potential. All of
the r -f voltage developed across the crystal
is available for excitation to the next stage,
but still is low for reasonable values of crystal current. For best operation a tube with low
heater - cathode capacitance is required. Excitation for the next stage may also be taken
from the cathode when using this circuit.
The circuit shown in fig ure 7A is also one used by
Pierce, but is more widely
referred to as the " \filler "oscillator. To avoid

Tuned -Plate

Crystal Oscillator

245

confusion, we shall refer to it as the tuned plate crystal oscillator. It is essentially an
Armstrong or tuned plate -tuned grid oscillator
with the crystal replacing the usual L -C grid
tank. The plate tank must be tuned to a frequency slightly higher than the anti -resonant

(parallel resonant) frequency of the crystal.
Whereas the Pierce circuits of figure 6 will
oscillate at (or very close to) the anti -resonant frequency of the crystal, the circuits of
figure 7 will oscillate at a frequency a little
above the anti -resonant frequency of the
crystal.
The diagram shown in figure 7A is the basic
circuit. The most popular version of the tuned plate oscillator employs a pentode or beam
tetrode with cathode bias to prevent excessive
plate dissipation when the circuit is not oscillating. The cathode resistor is optional. Its
omission will reduce both crystal current and
oscillator efficiency, resulting in somewhat
more output for a given crystal current. The
tube usually is an audio or video beam pentode
or tetrode, the plate -grid capacitance of such
tubes being sufficient to ensure stable oscillation but not so high as to offer excessive feedback with resulting high crystal current. The
6AG7 makes an excellent all- around tube for
this type circuit.

The usual type of crystal controlled h -f transmitter
Oscillator Circuits operates, at least part of
the time, on a frequency
which is an integral multiple of the operating
frequency of the controlling crystal. Hence,
oscillator circuits which are capable of providing output on the crystal frequency if desired, but which also can deliver output energy
on harmonics of the crystal frequency have
come into wide use. Four such circuits which
have found wide application are illustrated in
figures 7C, 7D, 7E, and 7F.
The circuit shown in figure 7C is recommended for use with harmonic -cut crystals
when output is desired on a multiple of the
oscillating frequency of the crystal. As an
example, a 25-Mc. harmonic-cut crystal may
be used in this circuit to obtain output on 50
Mc., or a 48 -Mc. harmonic-cut crystal may be
used to obtain output on the 144-Mc. amateur
band. The circuit is not recommended for use
with the normal type of fundamental- frequency
crystal since more output with fewer variable
elements can be obtained with the circuits of
7D and 7F.
The Pierce -harmonic circuit shown in figure 7D is satisfactory for many applications
which require very low crystal current, but
has the disadvantage that both sides of the
crystal are above ground potential. The Tri -tet
circuit of figure 7E is widely used and can
Pentode

Harmonic Crystal

246

Generation of
6J5,Erc

F

R

-F

6V6. 6AQ5,

RADIO

THE

Energy

6 AG7, 6AQ5.

ETC

3F

5763

002
150V.

+250

BASIC TUNED -PLATE OSCILLATOR

6AG7

F,

2F, 3F

+250 V.

V.

SPECIAL C RCUIT FOR USE WITH
HARMONIC CUT CRYSTAL.

RECOMMENDED TUNED -PLATE

OSCILLATOR

6AG7

F,

2F, 3F,

6AG7

4F

F.

2F, 3F,

4F

10111JF

150 ULF

22

+250

K

PIERCE HARMONIC CIRCUIT

"TRITET" CIRCUIT

Figure

V.

COLPITTS HARMONIC OSCILLATOR

7

COMMONLY USED CRYSTAL OSCILLATOR CIRCUITS
Shown at (A) is the basic tuned -plate crystal oscillator with a triode oscillator tube.
The plate tank must be tuned on the low -capacitance side of resonance to sustain
oscillation. (8) shows the tuned -plat oscillator as it is normally used with an a -f
power pentode to permit high output with relatively low crystal current.
Schematics (C), (D), (E), and (F) illustrate crystal oscillator circuits which can deliver moderate output energy on harmonics of the oscillating frequency of the crystal. (C) shows a special circuit which will permit use of a harmonic -cut crystal to
obtain output energy well into the v -h-f range. (D) is valuable when extremely low
crystal current is a requirement, but delivers relatively low output. (E) is commonly
used, but is subject to crystal damage if the cathode circuit is mistuned. (F) is
recommended as the most generally satisfactory from the standpoints of: low crystal current regardless of mis- adjustment, good output on harmonic frequencies, one
side of crystal is grounded, will oscillate with crystals from 1.5 to 70 Mc. without
adjustment, output tank may be tuned to the crystal frequency for fundamental output without stopping oscillation or changing frequency.

give excellent output with low crystal current.
However, the circuit has the disadvantages of
requiring a cathode coil, of requiring careful
setting of the variable cathode capacitor to
avoid damaging the crystal when changing frequency ranges, and of having both sides of
the crystal above ground potential.
The Colpitts harmonic oscillator of figure
7F is recommended as being the most generally satisfactory harmonic crystal oscillator
circuit since it has the following advantages:
(1) the circuit will oscillate with crystals
over a very wide frequency range with no
change other than plugging in or switching in
the desired crystal; (2) crystal current is ex-

tremely low; (3) one side of the crystal is
grounded, which facilitates crystal- switching
circuits; (4) the circuit will operate straight
through without frequency pulling, or it may
be operated with output on the second, third,
or fourth harmonic of the crystal frequency.
Crystal Oscillator
Tuning

The tunable circuits of all

oscillators

illustrated

should be tuned for maximum output as indicated by maximum excitation to the following stage, except that the
oscillator tank of tuned -plate oscillators (figure 7A and figure 7B) should be backed off
slightly towards the low capacitance side from

www.americanradiohistory.com

HANDBOOK

All -band Crystal Oscillator

6AG7

77. a 305
eaW M /N/JUC TOR
(2 0 vN)

+300

V.

NOTES

f"

I. Li'/sUN (2
OF eew 0301s)
(/" OF 80W .3003)
3. FOR 160 METER OPERATION ADD s 4/1/F. CONDENSER
BETWEEN PINS 418 OF 6A67. PLATE COIL= 35 2/P+.

2. L2 = /.gLN

(21-.0; Dew R awe)

4.

X' 7 MC.

CRYSTAL FOR HARMON /C OPERATION

Figure

8

ALL -3AND 6AG7 CRYSTAL OSCILLATOR
CAPABLE OF DRIVING
BEAM PENTODE TUBE

maximum output, as the oscillator then is in
a more stable condition and sure to start immediately when power is applied. This is especially important when the oscillator is keyed,
as for break-in c -w operation.

It is desirable to keep stray
shunt capacitances in the
crystal circuit as low as possible, regardless
of the oscillator circuit. If a selector switch
is used, this means that both switch and crystal sockets must be placed close to the oscillator tube socket. This is especially true of
harmonic -cut crystals operating on a comparatively high frequency. In fact, on the highest
frequency crystals it is preferable to use a
turret arrangement for switching, as the stray
capacitances can be kept lower.
Crystal Switching

Crystol Oscillator When the crystal

oscillator
is keyed, it is necessary
that crystal activity and oscillator -tube transconductance be moderately
high, and that oscillator loading and crystal
shunt capacitance be low. Below 2500 kc. and
above 6 Mc. these considerations become especially important. Keying of the plate voltage
(in the negative lead) of a crystal oscillator,
with the screen voltage regulated at about
150 volts, has been found to give satisfactory
results.
Keying

Versatile 6AG7
Crystal Oscillator
A

The 6AG7 tube may be
used in a modified Tri -tet

crystal oscillator, capable

of delivering sufficient power on all bands

247

from 160 meters through 10 meters to fully
drive a pentode tube, such as the 807, 2E26
or 6146. Such an oscillator is extremely useful for portable or mobile work, since it combines all essential exciter functions in one
tube. The circuit of this oscillator is shown
in figure 8. For 160, 80 and 40 meter operation the 6AG7 functions as a tuned-plate oscillator. Fundamental frequency crystals are
used on these three bands. For 20, 15 and 10
meter operation the 6AG7 functions as a Tritet oscillator with a fixed -tuned cathode circuit. The impedance of this cathode circuit
does not affect operation of the 6AG7 on the
lower frequency bands so it is left in the circuit at all times. A 7 -Mc. crystal is used for
fundamental output on 40 meters, and for harmonic output on 20, 15 and 10 meters. Crystal
current is extremely low regardless of the output frequency of the oscillator. The plate circuit of the 6AG7 is capable of tuning a frequency range of 2:1, requiring only two output
coils: one for 80-40 meter operation, and one
for 20, 15 and 10 meter operation. In some
cases it may be necessary to add 5 micromicrofarads of external feedback capacity between
the plate and control grid of the 6AG7 tube to
sustain oscillation with sluggish 160 meter

crystals.

Triode Overtone

Oscillators

The recent development of

reliable overtone crystals
capable of operation on the
third, fifth, seventh (or higher) overtones has
made possible v -h -f output from a low frequency crystal by the use of a double triode regenerative oscillator circuit. Some of the new
twin triode tubes such as the 12AU7, 12AV7
and 6J6 are especially satisfactory when used
in this type of circuit. Crystals that are ground
for overtone service may be made to oscillate
on odd overtone frequencies other than the
one marked on the crystal holder. A 24 -Mc.
overtone crystal, for example, is a specially
ground 8 -Mc. crystal operating on its third
overtone. In the proper circuit it may be made
to oscillate on 40 Mc. (fifth overtone), 56 Mc.
(seventh overtone), or 72 Mc. (ninth overtone).
Even the ordinary 8 -Mc. crystals not designed
for overtone operation may be made to oscillate readily on 24 Mc. (third overtone) in these
circuits.
A variety of overtone oscillator circuits is
shown in figure 9. The oscillator of figure 9A
is attributed to Frank Jones, W6AJF. The first
section of the 6J6 dual triode comprises a regenerative oscillator, with output on either the
third or fifth overtone of the crystal frequency.
The regenerative loop of this oscillator consists of a condenser bridge made up of C, and
C2, with the ratio C2 /C, determining the amount
of regenerative feedback in the circuit. With

www.americanradiohistory.com

248

Generation of

R

RADIO

THE

Energy

-F

2F 6J6
4F

6,9,10 o915 F

FOR 7 MC. CRYSTAL

+300

+300

AO

LI =28T e 24 ON NATIONAL

V

FORM
L 2. $ T Il
FORM

V.

©

JONES HARMONIC OSCILLATOR
12AÚ7

XPSO

ON NATIONAL XRSO

COLPITTS HARMONIC OSCILLATOR

6J6

3F

6, 9F

3F

/!

/s0

50

6,9F

RFC

FOR

+300

©

v.

L I= 9 7.

L2. 4

T

MC. CRYSTAL

a

5003 9iW M /N /DUCTOR
13003 817 MIN/DOCTOR

L

+300V

I

=

FOR 8MC. CRYSTAL
M NIDUC TOR,
TAP AT 37. FROM GRID ENO

/07 a 30 /I 64W

THESE COLS MADE FROM SINGLE SECT /ON
OF
NE TRN BROKEN TO
O /V /OE IINDUCTOR INTO TWO COILS S

REGENERATIVE HARMONIC OSCILLATOR

OD

REGENERATIVE HARMONIC OSCILLATOR

- 12AT7 (or 6AB4)

12AT7

6,9F

_

_

!00

F=IA4MC.

1_

L2
+200

LI=STe/E,
L2=/ T.
+300
E©

V.

f-

O.SPACED
NOOXUPWIRE,
O.

V.

CATHODE FOLLOWER OVERTONE OSCILLATOR

VARIOUS TYPES

OF

OVERTONE

OF

V.H.F. OVERTONE OSCILLATOR

Figure 9
OSCILLATORS USING
TUBES

an 8 -plc. crystal, output from the first section
of the 6J6 tube may be obtained on either 24
Mc. or 40 Mc., depending upon the resonant
frequency of the plate circuit inductor, L,. The
second half of the 636 acts as a frequency
multiplier, its plate circuit, L2, tuned to the
sixth or ninth harmonic frequency when L, is
tuned to the third overtone, or to the tenth
harmonic frequency when L, is tuned to the

fifth overtone.
Figure 9B illustrates a Colpitts overtone
oscillator employing a 636 tube. This is an
outgrowth of the Colpitts harmonic oscillator
of figure 7F. The regenerative loop in this

MINIATURE

case consists of

C

DOUBLE -TRIODE

C2 and RFC between the
grid, cathode and ground of the first section
of the 6J6. The plate circuit of the first section is tuned to the second overtone of the
crystal, and the second section of the 636
doubles to the fourth harmonic of the crystal.
This circuit is useful in obtaining 28 -Mc. output from a 7 -Mc. crystal and is highly popular
in mobile work.
The circuit of figure 9C shows a typical regenerative overtone oscillator employing a
12AÚ7 double triode tube. Feedback is controlled by the number of turns in L2, and the
coupling between L2 and L,. Only enough feed-

www.americanradiohistory.com

HANDBOOK

R

back should be employed to maintain proper
oscillation of the crystal. Excessive feedback
will cause the first section of the 12ÁU7 to

oscillate as

a

self- excited TNT oscillator,

independent of the crystal. A variety of this
circuit is shown in figure 9D, wherein a tapped
coil,
is used in place of the two separate
coils. Operation of the circuit is the same in
either case, regeneration now being controlled
by the placement of the tap on L,.
A cathode follower overtone oscillator is
shown in figure 9E. The cathode coil,
is
chosen so as to resonate with the crystal and
tube capacities just below the third overtone
frequency of the crystal. For example, with an
8 -Mc. crystal, L3 is tuned to 24 Mc.. L, resonates with the circuit capacities to 23.5 Mc.,
and the harmonic tank circuit of the second
section of the 12AT7 is tuned either to 48 Mc.
or 72 11c. If a 24 -Mc. overtone crystal is used
in this circuit, L, may be tuned to 72 Mc., L,
resonates with the circuit capacities to 70
Mc., and the harmonic tank circuit,
is tuned
to 144 Mc. If there is any tendency towards
self-oscillation in the circuit, it may be eliminated by a small amount of inductive coupling
between L2 and L3. Placing these coils near
each other, with the winding of L, correctly
polarized with respect to L3 will prevent selfoscillation of the circuit.
The use of a 144 -Mc. overtone crystal is
illustrated in figure 9F. A 6AB4 or one -half
of a 12AT7 tube may be used, with output
directly in the 2 -meter amateur band. A slight
amount of regeneration is provided by the one
which is loosely coupled to the
turn link,
144 -Mc. tuned tank circuit, L, in the plate circuit of the oscillator tube. If a 12AT7 tube
and a 110 -Mc. crystal are employed, direct output in the 220 -Mc. amateur band may be obtained from the second half of the 12AT7.

L

L

L

L

13 -4

Radio Frequency

Amplifiers
The output of the oscillator stage in a transmitter (whether it be self-controlled or crystal
controlled) must be kept down to a fairly low
level to maintain stability and to maintain a
factor of safety from fracture of the crystal
when one is used. The low power output of
the oscillator is brought up to the desired
power level by means of radio -frequency amplifiers. The two classes of r -f amplifiers that
find widest application in radio transmitters
are the Class B and Class C types.
The Class B

Amplifier

Class B amplifiers are used in a
radio -telegraph transmitter when
maximum power gain and mini-

-F

Amplifiers

249

output is desired in a particular stage. A Class B amplifier operates with
cutoff bias and a comparatively small amount
of excitation. Power gains of 20 to 200 or so
are obtainable in a well- designed Class B
amplifier. The plate efficiency of a Class B
c -w amplifier will run around 65 per cent.
mum harmonic

Another type of Class B ampli fier is the Class B linear stage
as employed in radiophone work.
This type of amplifier is used to increase the
level of a modulated carrier wave, and depends for its operation upon the linear relation between excitation voltage and output
voltage. Or, to state the fact in another manner, the power output of a Class B linear stage
varies linearly with the square of the excitation voltage.
The Class B linear amplifier is operated
with cutoff bias and a small value of excitation, the actual value of exciting power being
such that the power output under carrier conditions is one -fourth of the peak power capabilities of the stage. Class B linears are very
widely employed in broadcast and commercial
installations, but are comparatively uncommon
in amateur application, since tubes with high
plate dissipation are required for moderate
output. The carrier efficiency of such an amplifier will vary from approximately 30 per
cent to 35 per cent.
The Class B

Linear

Class C amplifiers are very wide ly used in all types of transmitters. Good power gain may be
obtained (values of gain from 3 to 20 are common) and the plate circuit efficiency may be,
under certain conditions, as high as 85 per
cent. Class C amplifiers operate with considerably more than cutoff bias and ordinarily with
a large amount of excitation as compared to a
Class B amplifier. The bias for a normal Class
C amplifier is such that plate current on the
stage flows for approximately 120° of the 360°
excitation cycle. Class C amplifiers are used
in transmitters where a fairly large amount of
excitation power is available and good plate
circuit efficiency is desired.

The Class C

Amplifier

The characteristic of a Class
C amplifier which makes it
linear with respect to
changes in plate voltage is that which allows
such an amplifier to be plate modulated for
radiotelephony. Through the use of higher bias
than is required for a c -w Class C amplifier
and greater excitation, the linearity of such
an amplifier may be extended from zero plate
voltage to twice the normal value. The output
power of a Class C amplifier, adjusted for
plate modulation, varies with the square of the

Plate Modulated
Class C

www.americanradiohistory.com

Generation

250

of

R

-F

THE

Energy

plate voltage. This is the same condition that
would take place if a resistor equal to the
voltage on the amplifier, divided by its plate
current, were substituted for the amplifier.
Therefore, the stage presents a resistive load

RADIO

Excessive grid current damages tubes by
overheating the grid structure; beyond a certain point of grid drive, no increase in power
output can be obtained for a given plate voltage.

to the modulator.
Grid Modulated

If the grid current to a

Class

amplifier is reduced to a
low value, and the plate loading is increased to the point where the plate
dissipation approaches the rated value, such
an amplifier may be grid modulated for radiotelephony. If the plate voltage is raised to
quite a high value and the stage is adjusted
carefully, efficiencies as high as 40 to 43 per
cent with good modulation capability and comparatively low distortion may be obtained.
Fixed bias is required. This type of operation
is termed Class C grid-bias modulation.
Class

Grid Excitation

Adequate grid

excitation

must be available for Class
service. The excitation for a
plate- modulated Class C stage must be sufficient to produce a normal value of d -c grid current with rated bias voltage. The bias voltage
preferably should be obtained from a combination of grid leak and fixed C -bias supply.
Cutoff bias can be calculated by dividing
the amplification factor of the tube into the
d-c plate voltage. This is the value normally
used for Class B amplifiers (fixed bias, no
grid resistor). Class C amplifiers use from 1
to 5 times this value, depending upon the available grid drive, or excitation, and the desired
plate efficiency. Less grid excitation is needed for c -w operation, and the values of fixed
bias (if greater than cutoff) may be reduced, or
the value of the grid leak resistor can be lowered until normal rated d-c grid current flows.
The values of grid excitation listed for each
type of tube may be reduced by as much as
50 per cent if only moderate power output and
plate efficiency are desired. When consulting
the tube tables, it is well to remember that
the power lost in the tuned circuits must be
taken into consideration when calculating the
available grid drive. At very high frequencies,
the r -f circuit losses may even exceed the
power required for actual grid excitation.
Link coupling between stages, particularly
to the final amplifier grid circuit, normally will
provide more grid drive than can be obtained
from other coupling systems. The number of
turns in the coupling link, and the location of
the turns on the coil, can be varied with respect to the tuned circuits to obtain the greatest grid drive for allowable values of buffer
or doubler plate current. Slight readjustments
sometimes can be made after plate voltage
has been applied to the driver tube.
B or

Class

Neutralization of
R.F. Amplifiers

13 -5

C

C

C

The plate -to -grid feedback capacitance of
triodes makes it necessary that they be neutralized for operation as r -f amplifiers at frequencies above about 500 kc. Those screen grid tubes, pentodes, and beam tetrodes which
have a plate -to -grid capacitance of 0.1 µµEd.
or less may be operated as an amplifier without neutralization in a well -designed amplifier
up to 30 Mc.
Neutralizing
Circuits

The object of neutralization is
to cancel or neutralize the capacitive feedback of energy from

plate to grid. There are two general methods
by which this energy feedback may be eliminated: the first, and the most common method,
is through the use of a capacitance bridge,
and the second method is through the use of a
parallel reactance of equal and opposite polarity to the grid -to-plate capacitance, to nullify the effect of this capacitance.
Examples of the first method are shown in
figure 10. Figure l0A shows a capacity neutralized stage employing a balanced tank circuit. Phase reversal in the tank circuit is obtained by grounding the center of the tank coil
to radio frequency energy by condenser C.
Points A and B are 180 degrees out of phase
with each other, and the correct amount of out
of phase energy is coupled through the neutralizing condenser NC to the grid circuit of
the tube. The equivalent bridge circuit of this
is shown in figure 11A. It is seen that the
bridge is not in balance, since the plate -filament capacity of the tube forms one leg of the
bridge, and there is no corresponding capacity
from the neutralizing condenser (point B) to
ground to obtain a complete balance. In addition, it is mechanically difficult to obtain a
perfect electrical balance in the tank coil, and
the potential between point A and ground and
point B and ground in most cases is unequal.
This circuit, therefore, holds neutralization
over a very small operating range and unless
tubes of low interelectrode capacity are used
the inherent unbalance of the circuit will permit only approximate neutralization.
Split-Stator
Plate Neutralization

www.americanradiohistory.com

Figure 10B shows the neu-

tralization circuit which is
most widely used in single ended r -f stages. The use of

HANDBOOK

Neutralization

Figure
COMMON

10

NEUTRALIZING CIRCUITS FOR SINGLE -ENDED AMPLIFIERS

split -stator plate capacitor makes the electrical balance of the circuit substantially independent of the mutual coupling within the coil
and also makes the balance independent of the
place where the coil is tapped. With conventional tubes this circuit will allow one neutralization adjustment to be made on, say, 28 Mc.,
and this adjustment usually will hold sufficiently close for operation on all lower frequency bands.
Condenser C, is used to balance out the
plate -filament capacity of the tube to allow a
perfect neutralizing balance at all frequencies.
The equivalent bridge circuit is shown in figure 11B. If the plate- filament capacity of the
tube is extremely low (100TH triode, for example), condenser C, may be omitted, or may
merely consist of the residual capacity of NC
to ground.
a

split grid tank circuit
also be used for neutralization of a triode tube as shown in figure
IOC. Out of phase voltage is developed across
a balanced grid circuit, and coupled through
NC to the single -ended plate circuit of the
tube. The equivalent bridge circuit is shown
in figure 11C. This circuit is in balance until
the stage is in operation when the loading effect of the tube upon one -half of the grid circuit throws the bridge circuit out of balance.
The amount of unbalance depends upon the
grid -plate capacity of the tube, and the amount
of mutual inductance between the two halves

Grid Neutralization

251

A

may

of the grid coil. If an r -f voltmeter is placed
between point A and ground, and a second
voltmeter placed between point B and ground
the loading effect of the tube will be noticeable. When the tube is supplied excitation
with no plate voltage, NC may be adjusted
until the circuit is in balance. When plate
voltage is applied to the stage, the voltage
from point A to ground will decrease, and the
voltage from point B to ground will increase,
both in direct proportion to the amount of circuit unbalance. The use of this circuit is not
recommended above 7 Mc., and it should be
used below that frequency only with low in-

ternal capacity tubes.
Two tubes of the same type
can be connected for push -pull
operation so as to obtain twice
as much output as that of a single tube. A
push -pull amplifier, such as that shown in figure 12 also has an advantage in that the circuit can more easily be balanced than a single tube r -f amplifier. The various inter -electrode
capacitances and the neutralizing capacitors
are connected in such a manner that the reactances on one side of the tuned circuits are
exactly equal to those on the opposite side.
For this reason, push -pull r -f amplifiers can
be more easily neutralized in very- high -frequency transmitters; also, they usually remain
in perfect neutralization when tuning the amplifier to different bands.
The circuit shown in figure 12 is perhaps
Push -Pull

Neutralization

Generation of

252

-F

R

THE

Energy

RADIO

F.AC

OA

BRIDGE EQUIVALENT OF FIGURE

IO -A

C

Figure 12
STANDARD CROSS- NEUTRALIZED
PUSH -PULL TRIODE AMPLIFIER

OB

BRIDGE EQUIVALENT OF FIGURE

10

-B

C

actance, coupling energy back from the plate
to the grid circuit. If this capacitance is paralleled with an inductance having the same
value of reactance of opposite sign, the reactance of one will cancel the reactance of
the other and a high -impedance tuned circuit

will result.
This neutralization circuit can be used on
ultra -high frequencies where other neutralization circuits are unsatisfactory. This is true
because the lead length in the neutralization
circuit is practically negligible. The circuit
can also be used with push -pull r -f amplifiers.
In this case, each tube will have its own neutralizing inductor connected from grid to plate.
The main advantage of this arrangement is
that it allows the use of single -ended tank
circuits with a single -ended amplifier.
The chief disadvantage of the shunt neutralized arrangement is that the stage must be reneutralized each time the stage is retuned to
a new frequency sufficiently removed that the
grid and plate tank circuits must be retuned to
resonance. However, by the use of plug -in
coils it is possible to change to a different
band of operation by changing the neutralizing coil at the same time that the grid and
plate coils are changed.
The 0.0001-pfd. capacitor in series with
the neutralizing coil is merely a blocking capacitor to isolate the plate voltage from the
grid circuit. The coil L will have to have a
very large number of turns for the band of operation in order to be resonant with the comparatively small grid -to -plate capacitance. But
since, in all ordinary cases with tubes operating on frequencies for which they were designed, the L/C ratio of the tuned circuit will
be very high, the coil can use comparatively
small wire, although it must be wound on air
or very low loss dielectric and must be insulated for the sum of the plate r-f voltage and
the grid r-f voltage.
from grid to plate

(RES.DUAL
CAPACITY)`

©

CG-r
(s1IALL)

RFC

BRIDGE EQUIVALENT OF FIGURE 10-G

Figure 11
EQUIVALENT NEUTRALIZING CIRCUITS

most commonly used arrangement for a
push -pull r -f amplifier stage. The rotor of the
grid capacitor is grounded, and the rotor of the
plate tank capacitor is by- passed to ground.
the

Shunt or Coil

Neutralization

The feedback of energy from
grid to plate in an unneutral-

ized r -f amplifier is

a

result of

the grid -to -plate capacitance of the amplifier
tube. A neutralization circuit is merely an
electrical arrangement for nullifying the effect
of this capacitance. All the previous neutralization circuits have made use of a bridge circuit for balancing out the grid -to -plate energy
feedback by feeding hack an equal amount of

energy of opposite phase.
Another method of eliminating the feedback
effect of this capacitance, and hence of neutralizing the amplifier stage, is shown in figure 13. The grid -to -plate capacitance in the
triode amplifier tube acts as a capacitive re-

www.americanradiohistory.com

Neutralizing

HANDBOOK

Procedure

253

grid excitation is applied, even though no primary a-c voltage is being fed to the plate transformer.

further check on the neutralization of any
amplifier can be made by noting whether
maximum grid current on the stage comes at
the same point of tuning on the plate tuning
capacitor as minimum plate current. This check
is made with plate voltage on the amplifier
and with normal antenna coupling. As the plate
tuning capacitor is detuned slightly from resonance on either side the grid current on the
A

r -f

Figure

13

COIL NEUTRALIZED AMPLIFIER
This neutralization circuit is very effective
with triode tubes on any frequency, but is

particularly effective in the v -h-f range. The
coil L is adjusted so that it resonates at the
operating frequency with the grid -to -plate
capacitance of the tube. Capacitor C may be
a very small unit of the low- capacitance
neutralizing type and is used to trim the circuit to resonance at the operating frequency.
If some means of varying the inductance of
the coil a small amount is available, the
trimmer capacitor is not needed.

stage should decrease the same amount and
without any sudden jumps on either side of
resonance. This will be found to be a very
precise indication of accurate neutralization
in either a triode or beam -tetrode r -f amplifier
stage, so long as the stage is feeding a load
which presents a resistive impedance at the
operating frequency.
Push-pull circuits usually can be more completely neutralized than single -ended circuits
at very high frequencies. In the intermediate
range of from 3 to 15 Mc., single -ended circuits will give satisfactory results.
Neutralization of
Screen -Grid

13 -6

R

-F

Amplifiers

Neutralizing
Procedure

An r-f amplifier is neutralized to prevent
self-oscillation or regeneration. A neon bulb,
a flashlight lamp and loop of wire, or an r -f
galvanometer can be used as a null indicator
for neutralizing low -power stages. The plate
voltage lead is disconnected from the r-f amplifier stage while it is being neutralized.
Normal grid drive then is applied to the r -f
stage, the neutralizing indicator is coupled
to the plate coil, and the plate tuning capacitor is tuned to resonance. The neutralizing

capacitor (or capacitors) then can be adjusted
until minimum r.f. is indicated for resonant
settings of both grid and plate tuning capacitors. Both neutralizing capacitors are adjusted simultaneously and to approximately the
same value of capacitance when a physically
symmetrical push -pull stage is being neutralized.
A final check for neutralization should be
made with a d-c milliammeter connected in the
grid leak or grid -bias circuit. There will be
no movement of the meter reading as the plate
circuit is tuned through resonance (without
plate voltage being applied) when the stage
is completely neutralized.
Plate voltage should be completely removed
by actually opening the d-c plate circuit. If
there is a d-c return through the plate supply,
a small amount of plate current will flow when

Radio-frequency amplifiers
using screen -grid tubes can
be operated without any ad-

ditional provision for neutralization at frequencies up to about 15 Mc.,
provided adequate shielding has been provided
between the input and output circuits. Special
v -h -f screen -grid and beam tetrode tubes such
as the 2E26, 807W, and 5516 in the low -power
category and HK -257B, 4E27/8001, 4 -125A,
and 4 -250A in the medium -power category can
frequently be operated at frequencies as high
as 100 Mc. without any additional provision
for neutralization. Tubes such as the 807,
2E22, HY -69, and 813 can be operated with
good circuit design at frequencies up to 30
Mc. without any additional provision for neutralization. The 815 tube has been found to
require neutralization in many cases above
20 Mc., although the 829B tube will operate
quite stably at 100 Mc. without neutralization.
None of these tubes, however, has perfect
shielding between the grid and the plate, a
condition brought about by the inherent inductance of the screen leads within the tube
itself. In addition, unless "watertight" shielding is used between the grid and plate circuits
of the tube a certain amount of external leakage between the two circuits is present. These
difficulties may not be serious enough to require neutralization of the stage to prevent
oscillation, but in many instances they show
up in terms of key -clicks when the stage in
question is keyed, or as parasitics when the
stage is modulated. Unless the designer of the
equipment can carefully check the tetrode

www.americanradiohistory.com

254

Generation of

R

-F

THE

Energy

Figure

RADIO

14

NEUTRALIZING CIRCUITS FOR
BEAM TETRODES

conventional cross neutralized circuit for use with push-pull beam tetrodes is
shown at (A). The neutralizing capacitors (NC) usually consist of small plates
or
rods mounted alongside the plate elements of the tubes. (B) and (C) show "grid
neutralized" circuits for use with a single -ended tetrode stage having either link
coupling or capacitive coupling into the grid tank. (D) shows a method of tuning the
screen -lead inductance to accomplish neutralization in a single -frequency -h-f
tetrode amplifier, while (E) shows a method of neutralization by increasing the vgrid
to -plate capacitance on a tetrode when the operating frequency is higher than
that
frequency where the tetrode is "self- neutralized" as a result of series resonance
in the screen lead. Methods (D) and (E) normally are not practicable at frequencies
below about 50 Mc. with the usual types of beam tetrode tubes.
A

stage for miscellaneous feedback between the
grid and plate circuits, and make the necessary circuit revisions to reduce this feedback
to an absolute minimum, it is wise to neutralize the tetrode just as if it were a triode tube.
In most push -pull tetrode amplifiers the simplest method of accomplishing neutralization
is to use the cross -neutralized capacitance
bridge arrangement as normally employed with
triode tubes. The neutralizing capacitances,
however, must be very much smaller than used
with triode tubes, values of the order of 0.2
wifd. normally being required with beam tetrode

tubes. This order of capacitance is far less
than can be obtained with a conventional neutralizing capacitor at minimum setting, so the
neutralizing arrangement is most commonly
made especially for the case at hand. Most
common procedure is to bring a conductor (connected to the opposite grid) in the vicinity of
the plate itself or of the plate tuning capacitor
of one of the tubes. Either one or two such
capacitors may be used, two being normally
used on a higher frequency amplifier in order
to maintain balance within the stage.
An example of this is shown in figure 14A.

www.americanradiohistory.com

H

Tetrode Neutralization

A N D B O O K

Neutralizing

single -ended tetrode r -f amplifier stage may be neutral ized in the same manner as
illustrated for a push -pull
stage in figure 14A, provided a split- stator
tank capacitor is in use in the plate circuit.
However, in the majority of single -ended tetrode r -f amplifier stages a single- section capacitor is used in the plate tank. Hence, other
neutralization procedures must be employed
when neutralization is found necessary.
The circuit shown in figure 14B is not a
true neutralizing circuit, in that the plate -togrid capacitance is not balanced out. However,
the circuit can afford the equivalent effect by
isolating the high resonant impedance of the
grid tank circuit from the energy fed back from
plate to grid. When NC and C are adjusted to
bear the following ratio to the grid -to-plate
capacitance and the total capacitance from
A

Single -Ended
Tetrode Stages

grid -to- ground in the output tube:
NC

CsP

C

Cas

both ends of the grid tank circuit will be at the
same voltage with respect to ground as a result
of r -f energy fed back to the grid circuit. This
means that the impedance from grid to ground
will be effectively equal to the reactance of
the grid -to- cathode capacitance in parallel
with the stray grid -to- ground capacitance, since
the high resonant impedance of the tuned circuit in the grid has been effectively isolated
from the feedback path. It is important to note
that the effective grid -to- ground capacitance
of the tube being neutralized includes the
rated grid -to- cathode or input capacitance of
the tube, the capacitance of the socket, wiring
capacitances and other strays, but it does not
include the capacitances associated with the
grid tuning capacitor. Also, if the tube is being excited by capacitive coupling from a preceding stage (as in figure 14C), the effective
grid-to- ground capacitance includes the output capacitance of the preceding stage and
its associated socket and wiring capacitances.

The provisions discussed in
the previous paragraphs are
for neutralization of the small,
though still important at the
higher frequencies, grid -to -plate capacitance
of beam -tetrode tubes. However, in the vicinity
of the upper- frequency limit of each tube type
the inductance of the screen lead of the tube
becomes of considerable importance. With a
tube operating at a frequency where the inductance of the screen lead is appreciable,
the screen will allow a considerable amount
of energy leak- through from plate to grid even
Cancellation of
Screen -Lead
Inductance

255

though the socket terminal on the tube is carefully by- passed to ground. This condition takes
place even though the socket pin is bypassed
since the reactance of the screen lead
will allow a moderate amount of r-f potential
to appear on the screen itself inside the electrode assembly in the tube. This effect has
been reduced to a very low amount in such
tubes as the Hytron 5516, and the Eimac 4X150A
and 4X500A but it is still quite appreciable in
most beam -tetrode tubes.
The effect of screen -lead inductance on the
stability of a stage can be eliminated at any
particular frequency by one of two methods.
These methods are: (1) Tuning out the screen lead inductance by series resonating the screen
lead inductance with a capacitor to ground.
This method is illustrated in figure 14D and is
commonly employed in commercially -built equipment for operation on a narrow frequency band
in the range above about 75 Mc. The other
method (2) is illustrated in figure 14E and
consists in feeding back additional energy
from plate to grid by means of a small capacitor connected between these two elements.
Note that this capacitor is connected in such
a manner as to increase the effective grid -toplate capacitance of the tube. This method
has been found to be effective with 807 tubes
in the range above 50 Mc. and with tubes such
as the 4 -125A and 4 -250A in the vicinity of
their upper frequency limits.
Note that both these methods of stabilizing
a beam-tetrode v -h -f amplifier stage by cancellation of screen -lead inductance are suitable only for operation over a relatively narrow
band of frequencies in the v -h-f range. At lower frequencies both these expedients for reducing the effects of screen -lead inductance
will tend to increase the tendency toward oscillation of the amplifier stage.

stage cannot be completely neutralized, the difficulty
usually can be traced to one or
more of the following causes: (1) Filament
leads not by- passed to the common ground of
that particular stage. (2) Ground lead from the
rotor connection of the split -stator tuning capacitor to filament open or too long. (3) Neutralizing capacitors in a field of excessive
r.f. from one of the tuning coils. (4) Electromagnetic coupling between grid and plate
coils, or between plate and preceding buffer
or oscillator circuits. (5) Insufficient shielding
or spacing between stages, or between grid
and plate circuits in compact transmitters.
(6) Shielding placed too close to plate circuit
coils, causing induced currents in the shields.
(7) Parasitic oscillations when plate voltage
is applied. The cure for the latter is mainly a
matter of cut and try -rearrange the parts,
Neutralizing

Problems

When a

-

256

Generation of

R

-F

THE

Energy

GRID
LEAK

RADIO

OUT

INTERWOUND COILS

(UNITY COUPLING)

Figure

CONVENTIONAL

16

TRIODE

FREQUENCY

MULTIPLIER
Figure

15

GROUNDED -GRID AMPLIFIER

This type of triode amplifier requires no
neutralization, but can be used only with
tubes having o relatively low plate -to- cathode
capacitance

change the length of grid or plate or neutralizing leads, insert a parasitic choke in the grid
lead or leads, or eliminate the grid r -f chokes
which may be the cause of a low- frequency

Small triodes such as the 604 operate satisfactorily as frequency multipliers, and can
deliver output well into the v -h -t ronge. Resistor R normally will have a value in the
vicinity of 100,000 ohms.

given output, because a moderate amount of
power is delivered to the amplifier load by the
driver stage of a grounded -grid amplifier.

13 -8

Frequency Multipliers

parasitic(in conjunction with plate r -f chokes).
13- 7

Grounded Grid

Amplifiers
Certain triodes have

grid configuration
results in very low
plate to filament capacitance when the control
grid is grounded, the grid acting as an effective shield much in the manner of the screen
in a screen -grid tube.
By connecting such a triode in the circuit of
figure 15, taking the usual precautions against
stray capacitive and inductive coupling between input and output leads and components,
a stable power amplifier is realized which requires no neutralization.
At ultra -high frequencies, where it is difficult to obtain satisfactory neutralization with
conventional triode circuits (particularly when
a wide band of frequencies is to be covered),
the grounded -grid arrangement is about the only
practicable means of employing a triode ama

and lead arrangement which

plifier.

Because of the large amount of degeneration
inherent in the circuit, considerably more excitation is required than if the same tube were
employed in a conventional grounded- cathode
circuit. The additional power required to drive
a triode in a grounded -grid amplifier is not
lost, however, as it shows up in the output circuit and adds to the power delivered to the
load. But nevertheless it means that a larger
driver stage is required for an amplifier of

Quartz crystals and variable- frequency oscillators are not ordinarily used for direct control of the output of high- frequency transmitters. Frequency multipliers are usually employed to multiply the frequency to the desired
value. These multipliers operate on exact multiples of the excitation frequency; a 3.6-Mc.
crystal oscillator can be made to control the
output of a transmitter on 7.2 or 14.4 Mc., or
on 28.8 Mc., by means of one or more frequency
multipliers. Chen used at twice frequency,
they are often termed frequency doublers. A
simple doubler circuit is shown in figure 16.
It consists of a vacuum tube with its plate circuit tuned to twice the frequency of the grid
driving circuit. This doubler can be excited
from a crystal oscillator or another multiplier
or amplifier stage.
Doubling is best accomplished by operating
the tube with high grid bias. The grid circuit
is driven approximately to the normal value of
d -c grid current through the r-f choke and grid leak resistor, shown in figure 16. The resistance value generally is from two to five times
as high as that used with the same tube for
straight amplification. Consequently, the grid
bias is several times as high for the same

value of grid current.

Neutralization is seldom necessary in a
doubler circuit, since the plate is tuned to
twice the frequency of the grid circuit. The
impedance of the grid driving circuit is very
low at the doubling frequency, and thus there
is little tendency for self -excited oscillation.

www.americanradiohistory.com

HANDBOOK

Frequency Multipliers

257

TANN CIRCUIT OUTPUT VOLTAGE

s

I

(CUTOrr)

n A A
\f/ \I

A

N

C,,

K

GI

I,J

N

\

L1

M

O

A

á

A

\\

U1

I

1

W

T

1

N(cuTOn)------¡--P--t-- i - - -1

\
\

i

EXCITATION
VOLTAGE

\O

Figure 18
ILLUSTRATING THE ACTION OF

A

FREQUENCY DOUBLER

degrees or less. Under these conditions
the efficiency will be on the same order as
the reciprocal of the harmonic on which the
stage operates. In other words the efficiency
of a doubler will be approximately % or 50
per cent, the efficiency of a tripler will be
approximately / or 33 per cent and that of a
quadrupler will be about 25 per cent. With good
stage design the efficiency can be somewhat
greater than these values, but as the angle
of flow is made greater than these limiting
values, the efficiency falls off rapidly. The
reason is apparent from a study of figure 18.
The pulses ABC, EFG, JKL illustrate 180 degree excitation pulses under Class B operation, the solid straight line indicating cutoff
bias. If the bias is increased by N times, to
the value indicated by the dotted straight line,
and the excitation increased until the peak
r -f voltage with respect to ground is the same
as before, then the excitation frequency can
be cut in half and the effective excitation
pulses will have almost the same shape as
before. The only difference is that every other
pulse is missing; MNO simply shows where
the missing pulse would go. However, if the
Q of the plate tank circuit is high, it will have
sufficient flywheel effect to carry over through
the missing pulse, and the only effect will be
that the plate input and r -f output at optimum
loading drop to approximately half. As the input frequency is half the output frequency, an
efficient frequency doubler is the result.
By the same token, a tripler or quadrupler
can be analyzed, the tripler skipping two excitation pulses and the quadrupler three. In
each case the excitation pulse ideally should
be short enough that it does not exceed 180
degrees at the output frequency; otherwise the
excitation actually is bucking the output over
a portion of the cycle.
In actual practice, it is found uneconomical
to provide sufficient excitation to run a tripler
or quadrupler in this fashion. Usually the ex45

Figure 17
FREQUENCY MULTIPLIER CIRCUITS
The output of a triode v-h -f frequency multiplier often maybe increased by neutralization
of the grid-to -plate capacitance as shown at
(A) above. Such o stage also may be operated as a straight amplifier when the occasion demands. A pentode frequency multiplier is shown at (B). Conventional power
tetrodes operate satisfactorily as multipliers
so long as the output frequency is below
about 100 Mc. Above this frequency special
v -h -f tetrodes must be used to obtain satisfactory output.

Frequency doublers require bias of several
times cutoff; high -ft tubes therefore are desirable for this type of service. Tubes which
have amplification factors from 20 to 200 are
suitable for doubler circuits. Tetrodes and
pentodes make excellent doublers. Low -ft
triodes, having amplification constants of from
3 to 10, are not applicable for doubler service.
In extreme cases the grid voltage must be as
high as the plate voltage for efficient doubling
action.
Angle of Flow
in Frequency

The angle of plate current flow
in a frequency multiplier is a
Multipliers
very important factor in determining the efficiency. As the
angle of flow is decreased for a given value
of grid current, the efficiency increases. To
reduce the angle of flow, higher grid bias is
required so that the grid excitation voltage
will exceed the cutoff value for a shorter portion of the exciting -voltage cycle. For a high
order of efficiency, frequency doublers should
have an angle of flow of 90 degrees or less,
tripiers 60 degrees or less, and quadruplers

www.americanradiohistory.com

Generation of

258

Figure

R

-F

THE

Energy

RADIO

19

Figure

PUSH -PUSH

FREQUENCY DOUBLER
The output of o doubler stage may be materially increased through the use of a push -push
circuit such as illustrated above.

citation pulses will be at least 90 degrees at
the exciting frequency, with correspondingly
low efficiency, but it is more practicable to
accept the low efficiency and build up the output in succeeding amplifier stages. The efficiency can become quite low before the power
gain becomes less than unity.
Two tubes can be connected in
parallel to give twice the output
of a single -tube doubler. If the
grids are driven out of phase instead of in
phase, the tubes then no longer work simultaneously, but rather one at a time. The effect
is to fill in the missing pulses (figure 18).
Not only is the output doubled, but several
advantages accrue which cannot be obtained
by straight parallel operation.
Chief among these is the effective neutralization of the fundamental and all odd harmonics, an advantage when spurious emissions
must be minimized. Another advantage is that
when the available excitation is low and excitation pulses exceed 90 degrees, the output
and efficiency will be greater than for the
same tubes connected in parallel.
The same arrangement may be used as a
quadrupler, with considerably better efficiency
than for straight parallel operation, because
seldom is it practicable to supply sufficient
excitation to permit 45 degree excitation
pulses. As pointed out above, the push -push
arrangement exhibits better efficiency than a
single ended multiplier when excitation is inadequate for ideal multiplier operation.
A typical push -push doubler is illustrated
in figure 19. When high transconductance tubes
are employed, it is necessary to employ a
split- stator grid tank capacitor to prevent self
oscillation; with well screened tetrodes or
pentodes having medium values of transconductance, a split -coil arrangement with a single- section capacitor may be employed (the

20

PUSH -PULL

FREQUENCY TRIPLER
The push -pull tripler is advantageous in the
v -h-f ronge since circuit balance is maintained both in the input and output circuits.
If the circuit is neutralized it may be used

either as a straight amplifier or as a tripler.
Either triodes or tetrodes may be used; dual unit tetrodes such as the 815, 832A, and

8298 are particularly effective in the v-h -f
range.

center tap of the grid coil being by- passed to
ground).

Push -Push

Multipliers

Push -Pull Frequency

It is frequently desirable
in the case of u -h -f and
v -h -f transmitters
that
frequency multiplication stages be balanced
with respect to ground. Further it is just as
easy in most cases to multiply the crystal or
v -f -o frequency by powers of three rather than
multiplying by powers of two as is frequently
done on lower frequency transmitters. Hence
the use of push -pull tripiers has become quite
prevalent in both commercial and amateur
v -h -f and u -h -f transmitter designs. Such stages
are balanced with respect to ground and appear
in construction and on paper essentially the
same as a push -pull r -f amplifier stage with
the exception that the output tank circuit is
tuned to three times the frequency of the grid
tank circuit. A circuit for a push -pull tripler
stage is shown in figure 20.
A push -pull tripler stage has the further
advantage in amateur work that it can also be
used as a conventional push -pull r -f amplifier
merely by changing the grid and plate coils
so that they tune to the same frequency. This
is of some advantage in the case of operating
the 50 -Mc. band with 50 -bic. excitation, and
then changing the plate coil to tune to 144
Mc. for operation of the stage as a tripler from
excitation on 48 Mc. This circuit arrangement
is excellent for operation with push -pull beam
tetrodes such as the 6360 and 829B, although
a pair of tubes such as the 2E26, or 5763 could
just as well be used if proper attention were
given to the matter of screen -lead inductance.
Tripiers

www.americanradiohistory.com

Tank

HANDBOOK
Tank Circuit
Capacitances

13 -9

It is necessary that the proper value of Q
plate tank circuit of any r -f
amplifier. The following section has been devoted to a treatment of the subject, and charts
are given to assist the reader in the determination of the proper L/C ratio to be used in a
radio -frequency amplifier stage.
A Class C amplifier draws plate current in
the form of very distorted pulses of short duration. Such an amplifier is always operated into a tuned inductance- capacitance or tank circuit which tends to smooth out these pulses,
by its storage or tank action, into a sine wave

DYNAMIC
CHARACTERISTIC

Tank Circuit

Q

As stated before, the tank cir-

cuit of a Class C amplifier
receives energy in the form of short pulses of
plate current which flow in the amplifier tube.
But the tank circuit must be able to store
enough energy so that it can deliver a current
essentially sine wave in form to the load. The
ability of a tank to store energy in this manner may be designated as the effective Q of
the tank circuit. The effective circuit Q may
be stated in any of several ways, but essentially the Q of a tank circuit is the ratio of the
energy stored to 2e times the energy lost per
cycle. Further, the energy lost per cycle must,
by definition, be equal to the energy delivered
to the tank circuit by the Class C amplifier
tube or tubes.
The Q of a tank circuit at resonance is equal
to its parallel resonant impedance (the resonant impedance is resistive at resonance) divided by the reactance of either the capacitor or the inductor which go to make up the
tank. The inductive reactance is equal to the
capacitive reactance, by definition, at resonance. Hence we may state:

259

A_

(0\

be used in the

of radio -frequency output. Any wave -form distortion of the carrier frequency results in harmonic interference in higher- frequency channels.
A Class A r-f amplifier would produce a sine
wave of radio -frequency output if its exciting
waveform were also a sine wave. However, a
Class A amplifier stage converts its d -c input
to r -f output by acting as a variable resistance,
and therefore heats considerably. A Class C
amplifier when driven hard with short pulses
at the peak of the exciting waveform acts more
as an electronic switch, and therefore can convert its d -c input to r -f output with relatively
good efficiency. Values of plate circuit efficiency from 65 to 85 per cent are common in
Class C amplifiers operating under optimum
conditions of excitation, grid bias, and loading.

Circuits

GRID SWING

Figure

21

AMPLIFIER OPERATION
Plate current pulses are shown at (A), (e),
and (C). The dip in the top of the plate current waveform will occur when the excitation
voltage is such that the minimum plate voltage dips below the maximum grid voltage.
A detailed discussion of the operation of
Class C amplifiers is given in Chapter Seven.
CLASS

C

Q =

-=RL

RL

Xc

XL

where RL is the resonant impedance of the
tank and Xc is the reactance of the tank capacitor and XL is the reactance of the tank

coil. This value of resonant impedance, RL,
is the load which is presented to the Class C
amplifier tube in a single -ended circuit such
as shown in figure 21.

The value of load impedance, RL, which the
Class C amplifier tube sees may be obtained,
looking in the other direction from the tank
coil, from a knowledge of the operating conditions on the Class C tube. This load impedance may be obtained from the following expression, which is true in the general case of
any Class C amplifier:
Epm=

RL
2

Np lb Ebb

where the values in the equation have the characteristics listed in the beginningof Chapter 6.
The expression above is academic, since
the peak value of the fundamental component
of plate voltage swing, Epm, is not ordinarily
known unless a high -voltage peak a-c voltmeter
is available for checking. Also, the decimal
value of plate circuit efficiency is not ordinarily known with any degree of accuracy. However, in a normally operated Class C amplifier

Generation of

260

R

> s
w

>

!

54
V-

8 3.
u x

100

,S

10

TANK CIRCUIT

30

25

20
Q

THE

Energy

-F

which means simply that the resistance presented by the tank circuit to the Class C tube
is approximately equal to one -half the d -c load
resistance which the Class C stage presents
to the power supply (and also to the modulator
in case high -level modulation of the stage is
to be used).
Combining the above simplified expression
for the r -f impedance presented by the tank to
the tube, with the expression for tank Q given
in a previous paragraph we have the following
expression which relates the reactance of the
tank capacitor or coil to the d-c input to the

Class

C

stage:
XC

Figure 22

RELATIVE HARMONIC OUTPUT
PLOTTED AGAINST TANK CIRCUIT

.v

Rd. c.

\\\\1
\
\
II
2

20000
15

10

i
Ó
>

w
I-

XL

,

Rd.c.

The above expression is the basis of the
usual charts giving tank capacitance for the
various bands in terms of the d -c plate voltage
and current to the Class C stage, including
the charts of figure 23, figure 24 and figure 25.
Harmonic Rodio-

The problem of

harmonic

radiation from transmitters
has long been present, but
it has become critical only relatively recently
along with the extensive occupation of the
v -h -f range. Television signals are particularly
susceptible to interference from other signals
falling within the pass band of the receiver,
so that the TVI problem has received the major
emphasis of all the services in the v -h -f range
which are susceptible to interference from
harmonics of signals in the h -f or lower v -h -f
range.
tion vs.

Q

\11111III1N1
111111!\IIIIIII
\\
Q=12

II

w

=

2Q
Q

the plate voltage swing will be approximately
equal to 0.85 to 0.9 times the d -c plate voltage
on the stage, and the plate circuit efficiency
will be from 70 to 80 per cent (Np of 0.7 to
0.8), the higher values of efficiency normally
being associated with the higher values of
plate voltage swing. With these two assumptions as to the normal Class C amplifier, the
expression for the plate load impedance can
be greatly simplified to the following approximate but useful expression:
RL

RADIO

III

IIII
mum
111

.

11

1

NEUTRALIZING
COIL

\11'I
\\111Ii
1111, I\IIII
(I

RFC

-e

1110M1121111101111111111

3

10

20

IIII!ÍiNHO!i
30

100

500

200

TOTAL CAPACITANCE ACROSS LC

C

1000

2000

RCUIT (CO

O

Figure 23

PLATE -TANK CIRCUIT ARRANGEMENTS
Shown above in the case of each of the tank circuit types is the recommended tank circuit capacitance. (A) is a conventional tetrode amplifier, (B) is a coil -neutralized triode amplifier,
(C) is a grounded-grid triode amplifier, (D) is a grid -neutralized triode amplifier.

www.americanradiohistory.com

re

HANDBOOK

10

Tank

Circuits

261

\.,'..O..U...n
\/
\11 III
IIII
ÌIÌI!I\1iIÌCIÌI\\IIIIIÌ
MONIIIMMIIIMUM111

\\111i111\11111111111111

1MINE111011; I111111MII1,vIM11111MMnII
1.3

o
>

á
II

¢

1

11

1

1

41 1
11

1

1

.,

IlliliE1111P11 1111111

et

11111111111111111111

10
2 3
5 7
lo 30 50 100 lao
500 1000
CORRECT VALUES OF TANK CIRCUIT CAPACITANCE (C FOR
OPERATING Q OF 12 WITH SINGLE-ENDED SPLIT TANK COILS

Figure 24

PLATE -TANK CIRCUIT ARRANGEMENTS
Shown above for each of the tank circuit types is the recommended tank circuit capacitance of
the operating frequency for an operating Q of 12. (A) is a split -stator tank, each section of which
is twice the capacity value read on the graph. (8) is circuit using tapped coil for phase reversal.

Inspection of figure 22 will show quickly
that the tank circuit of a Class C amplifier
should have an operating Q of 12 or greater
to afford satisfactory rejection of second harmonic energy. The curve begins to straighten
out above a Q of about 15, so that a considerable increase in Q must be made before an appreciable reduction in second -harmonic energy
is obtained. Above a circuit Q of about 10 any
increase will not afford appreciable reduction
in the third -harmonic energy, so that additional
harmonic filtering circuits external to the amplifier proper must be used if increased attenuation of higher order harmonics is desired.
The curves also show that push -pull amplifiers
may be operated at Q values of 6 or so, since
the second harmonic is cancelled to a large
extent if there is no unbalanced coupling between the output tank circuit and the antenna
system.

Figures 23, 24 and 25 illustrate the correct value
of tank capacity for various circuit configurations. A Q value of 12
has been chosen as optimum for single ended
circuits, and a value of 6 has been chosen for
push -pull circuits. Figure 23 is used when a
single ended stage is employed, and the capacitance values given are for the total capacitance across the tank coil. This value includes the tube interelectrode capacitance
(plate to ground), coil distributed capacitance,
wiring capacities, and the value of any lowCapacity Charts for

Correct Tank

Q

inductance plate -to- ground by -pass capacitor
as used for reducing harmonic generation, in
addition to the actual "in -use" capacitance
of the plate tuning capacitor. Total circuit
stray capacitance may vary from perhaps 5
micromicrofarads for a v -h -f stage to 30 micro microfarads for a medium power tetrode h -f
stage.
When a split plate tank coil is employed in
the stage in question, the graph of figure 24
should be used. The capacity read from the
graph is the total capacity across the tank
coil. If the split- stator tuning capacitor is
used, each section of the capacitor should
have a value of capacity equal to twice the
value indicated by the graph. As in the case
of figure 23, the values of capacity read on
the graph of figure 24 include all residual circuit capacities.
For push-pull operation, the correct values
of tank circuit capacity may be determined
with the aid of figure 25. The capacity values
obtained from figure 25 are the effective values
across the tank circuit, and if a split- stator
tuning capacitor is used, each section of the
capacitor should have a value of capacity equal to twice the value indicated by the graph.
As in the case of figures 23 and 24, the values
of capacity read on the graph of figure 25 include all residual circuit capacities.
The tank circuit operates in the same manner whether the tube feeding it is a pentode,
beam tetrode, neutralized triode, grounded grid triode, whether it is single ended or push-

www.americanradiohistory.com

Generation of

262

R

-F

THE

Energy

RADIO

20000

Q°6
16000

0
8000

uz

á
>ú
«

6000

0:

w

J

O. a.

ú

é

1000
2

3

!

7

10

20

30

30

00

200

300 1000
) FOR

CORRECT VALUES OF TANK CIRCUIT CAPACITANCE C
OPERAT NG Q OF 6 WITH PUSH -PULL TANK CIRCUITS

Figure 25

PLATE -TANK CIRCUIT ARRANGEMENTS FOR PUSH -PULL STAGES
Shown above is recommended tank circuit capacity at operating frequency for a Q of 6. (A) is
split-stator tank, each section of which is twice the capacity value read on the graph. (B) is
circuit using topped coil for phase reversal.

pull, or whether it is shunt fed or series fed.
The important thing in establishing the operating Q of the tank circuit is the ratio of the
loaded resonant impedance across its terminals to the reactance of the L and the C which
make up the tank.
Due to the unknowns involved in determining circuit stray capacitances it is sometimes
more convenient to determine the value of L
required for the proper circuit Q (by the method
discussed earlier in this Section) and then to
vary the tuned circuit capacitance until resonance is reached. This method is most frequently used in obtaining proper circuit Q in
commercial transmitters.
The values of Rp for using the charts are
easily calculated by dividing the d -c plate supply voltage by the total d -c plate current (expressed in amperes). Correct values of total
tuning capacitance are shown in the chart for
the different amateur bands. The shunt stray
capacitance can be estimated closely enough
for all practical purposes. The coil inductance
should then be chosen which will produce
resonance at the desired frequency with the
total calculated tuning capacitance.
The Q of a circuit depends
upon the resistance in series
with the capacitance and inductance. This series resistance is very low
for a low -loss coil not loaded by an antenna
circuit. The value of Q may be from 100 to 600
under these conditions. Coupling an antenna
Effect of Loading on

Q

circuit has the effect of increasing the series
resistance, though in this case the power is
consumed as useful radiation by the antenna.
Mathematically, the antenna increases the
value of R in the expression Q = oiL /R where
L is the coil inductance in microhenrys and
is the term 2nf, f being in megacycles.
The coupling from the final tank circuit to
the antenna or antenna transmission line can
be varied to obtain values of Q from perhaps
3 at maximum coupling to a value of Q equal
to the unloaded Q of the circuit at zero antenna coupling. This value of unloaded Q can
be as high as 500 or 600, as mentioned in the
preceding paragraph. However, the value of
Q = 12 will not be obtained at values of normal d -c plate current in the Class C amplifier
stage unless the C -to -L ratio in the tank circuit is correct for that frequency of operation.
To determine the required
tuning capacitor air gap for
a particular amplifier circuit it is first necessary to estimate the peak
r-f voltage which will appear between the
plates of the tuning capacitor. Then, using
figure 26, it is possible to estimate the plate
spacing which will be required.
The instantaneous r -f voltage in the plate
circuit of a Class C amplifier tube varies from
nearly zero to nearly twice the d -c plate voltage. If the d -c voltage is being 100 per cent
modulated by an audio voltage, the r-f peaks
will reach nearly four times the d -c voltage.

Tuning Capacitor
Air Gap

www.americanradiohistory.com

HANDBOOK

L

and

Pi

Networks

RP RA(Q2+1)(txACT)

FIGURE 26
USUAL BREAKDOWN RATINGS OF
COMMON PLATE SPACINGS
Air-gap in
Peak voltage
inches
breakdown
.030
1,000
.050
2,000
.070
3,000
.100
4,000
.125
4,500
.150
5,200
.170
6,000
.200
7,500
.250
9,000
.350
11,000
.500
15,000
.700
20,000

Recommended air -gap for use when no d -c
voltage appears across plate tank condenser
(when plate circuit is shunt fed, or when the
plate tank condenser is insulated from
ground).
D.C. PLATE

VOLTAGE

400
600

750
1000
1250
1500

2000
2500
3000
3500

263

C.W.

PLATE
MOD.

.030
.050
.050
.070

.050
.070
.084
.100

.070

RP

=

Q2 RA (APPROX.)

Q=xs__X.. -BL-B.e
RA

RA

XC

XL

XL =Xc

RF

+e

RP= APPROX. LATE VOLTAG4
'PLATE CURRENT
RP= 225 RA

=
FOR OPERATING
Q OF 15

CIRCUIT
XC

XL=

Figure
THE

*
19

27

NETWORK

IMPEDANCE
TRANSFORMER
The L network is useful with a moderate
operating Q for high values of impedance
transformation, and it may be used for applications other than in the plate circuit of a
tube with relatively low values of operating
Q for moderate impedance transformations.
Exact and approximate design equations ore
given.
L

.144

.078
.100
.175
.200
.250

.200
.250
.375
.500
.600

should be multiplied by 1.5 for
some safety factor when d -c voltage appears

Spacings

across plate tank condenser.

These rules apply to a loaded amplifier or
buffer stage. If either is operated without an
r -f load, the peak voltages will be greater and
can exceed the d-c plate supply voltage. For
this reason no amplifier should be operated
without load when anywhere near normal d -c
plate voltage is applied.
If a plate blocking condenser is used, it
must be rated to withstand the d -c plate voltage plus any audio voltage. This capacitor
should be rated at a d -c working voltage of at
least twice the d-c plate supply in a plate modulated amplifier, and at least equal to the d -c
supply in any other type of r -f amplifier.

between the plate tank circuit of an amplifier
and a transmission line, or they may be used
to match directly from the plate circuit of an
amplifier to the line without the requirement
for a tank circuit -provided the network is designed in such a manner that it has sufficient
operating Q for accomplishing harmonic attenuation.
The L Matching
Network

L and Pi Matching
Networks

The L network is of limited
utility in impedance matching since its ratio of impedance transformation is fixed at a value equal
to (Q2 +1). The operating Q may be relatively
low (perhaps 3 to 6) in a matching network between the plate tank circuit of an amplifier
and a transmission line; hence impedance
transformation ratios of 10 to 1 and even lower
may be attained. But when the network also
acts as the plate tank circuit of the amplifier
stage, as in figure 27, the operating Q should
be at least 12 and preferably 15. An operating
Q of 15 represents an impedance transformation of 225; this value normally will be too
high even for transforming from the 2000 to
10,000 ohm plate impedance of a Class C amplifier stage down to a 50 -ohm transmission

The L and pi networks often can be put to
advantageous use in accomplishing an impedance match between two differing impedances.
Common applications are the matching between
a transmission line and an antenna, or between
the plate circuit of a single -ended amplifier
stage and an antenna transmission line. Such
networks may be used to accomplish a match

However, the L network is interesting since
the basis of design for the pi network.
Inspection of figure 27 will show that the L
network in reality must be considered as a
parallel- resonant tank circuit in which RA
represents the coupled -in load resistance;
only in this case the load resistance is directly coupled into the tank circuit rather than
being inductively coupled as in the conven-

13 -10

line.

it forms

www.americanradiohistory.com

Generation of

264

R

-F

THE

Energy

RADIO

pacitance may be obtained for an operating Q
of 12 by reference to figures 23, 24 and 25.
The inductive arm in the pi network can be
thought of as consisting of two inductances
in series, as illustrated in figure 28. The first
is that value of
portion of this inductance,
inductance which would resonate with C, at
the operating frequency -the same as in a conventional tank circuit. However, the actual
value of inductance in this arm of the pi network, L10, will be greater than L, for normal
values of impedance transformation. For high
transformation ratios Lot will be only slightly
greater than Li; for a transformation ratio of
1.0, L10t will be twice as great as L. The
amount of inductance which must be added to
L, to restore resonance and maintain circuit
Q is obtained through use of the expression

L

Roc

Ebb

=

XCZ

I

Rp
XCn

-RA

Rp

RA(Q2+1)-Rp

Roc.

z
Rp
=

XCi
RI12tXG22

XLZ'

RAZ

ALTOT.

XL1+XL2

Q

XL,

Á

for X12 in figure 28.

Figure 28
THE PI NETWORK
The pi network is valuable for use as an impedance transformer over a wide ratio of
transformation values. The operating Q should
be at least 12 and preferably 15 to 20 when
the circuit is to be used in the plate circuit

of

Class C amplifier. Design equations
are given above. The inductor Ltot represents a single inductance, usually variable,
with a value equal to the sum of Lt and L2.
a

tional arrangement where the load circuit is
coupled to the tank circuit by means of a link.
When RA is shorted, L and C comprise a conventional parallel- resonant tank circuit, since
for proper operation L and C must be resonant
in order for the network to present a resistive
load to the Class C amplifier.

pi impedance matching
network, illustrated in figure
28, is much more general in its application
than the L network since it offers greater harmonic attenuation, and since it can be used
to match a relatively wide range of impedances
while still maintaining any desired operating
Q. The values of C, and L, in the pi network
of figure 28 can be thought of as having the
same values of the L network in figure 27 for
the same operating Q, but what is more important from the comparison standpoint these values will be the same as in a conventional tank
circuit.
The value of the capacitance may be determined by calculation, with the operating Q and
the load impedance which should be reflected
to the plate of the Class C amplifier as the
two knowns -or the actual values of the ca-

The Pi Network

The

The peak voltage rating of the main tuning
capacitor C, should be the normal value for a
Class C amplifier operating at the plate volt-

age to be employed. The inductor L101 may be
a plug -in coil which is changed for each band
of operation, or some sort of variable inductor
may be used. A continuously variable

slider -

type of variable inductor, such as used in certain items of surplus military equipment, may
be used to good advantage if available, or a
tapped inductor such as used in the ART -13
may be employed. However, to maintain good
circuit Q on the higher frequencies when a
variable or tapped coil is used on the lower
frequencies, the tapped or variable coil should
be removed from the circuit and replaced by
a smaller coil which has been especially designed for the higher frequency ranges.
The peak voltage rating of the output or
loading capacitor, C2i is determined by the
power level and the impedance to be fed. If a
50 -ohm coaxial line is to be fed from the pi
network, receiving -type capacitors will be
satisfactory even up to the power level of a
plate -modulated kilowatt amplifier. In any
event, the peak voltage which will be impressed across the output capacitor is expressed by: Epk2 = 2 R. Wo, where Epk is the
peak voltage across the capacitor, R. is the
value of resistive load which the network is
feeding, and W. is the maximum value of the
average power output of the stage. The harmonic attenuation of the pi network is quite
good, although an external low -pass filter will
be required to obtain harmonic attenuation
value upward of 100 db such as normally required. The attenuation to second harmonic
energy will be approximately 40 db for an operating Q of 15 for the pi network; the value
increases to about 45 db for a 1:1 transformation and falls to about 38 db for an impedance
step -down of 80:1, assuming that the operating Q is maintained at 15.

www.americanradiohistory.com

HANDBOOK

Grid

Bias

265

RFC
CB

COAX
OUTPUT

-B

E.
Zx[B

PLATE LOAD (OHMS)

WHERE ES IS PLATE VOLTAGE
AND I B IS PLATE CURRENT

IN

4MPE II

CS

Ce- .000252/F

MICA CAPACITOR RATED AT TWICE THE D.C.
PLATE VOLTAGE

RFC i -Ne 28 ENAMELED. CLOSE -WOUND ON

I'OIA.,

RFC2Est

mated

I.

n

1,500

2,000

2,500

3,000

3,500

4,000

4,500

5,000

7
14
21

360
180
90
60

210
105

180
90

120
60

110
56

52
35

45

34
23

30
20

28

65

45

26

33

155
76
38
25
19

135
68

28

280
140
70
47
35

17

15

4.5
2.2

3.5 Mc

in puf, 3.5 Mc
7
14
21
28

gal,

6.5
3.2
1.6

3.5 Mc
7
14
21

28

31

8.5
4.2

10.5
5.2

12.5
6.2

14

15.5

7

2.6

3.1

7.8
3.9
2.6

4.5

1.95

2.25

18
9

19
14

20
10
5

6,000

NOTES

actuel capacitance setting
45 for C, equals the value in this
23 table minus the published tube
15 output
capacitance. Air gap
11
approx. 10 mils 100 v E,,.
90

The

25

Inductance

values

are for

a

12.5 50 -ohm load. For a 70 -ohm
6.2 load, values ore approx. 3%

0.73
0.55

1.08

2.1
1.38

2.05

0.8

1.05

1.7
1.28

3.5
2.3

1.55

1.7

2,400
1,200
600
400
300

2,100
1,060
530
350
265

1,800
900

1,550
760

1,400
700

1,250
630

1,100
560

1,000
500

900
460

450
300
225

380
250
190

350
230
175

320
210
160

280

250

230

700 For 50 -ohm transmission line.
350 Air gap for Cr is approx. 1
175 mil 100 v E,.

185
140

165
125

155
115

120
90

1.800
900
450
300
225

1,500
750
370
250
185

1,300
650
320
215
160

1,100
560
280

1,000
500
250

900
450
220

800
400
200

720
360
180

640
320
160

190

170
125

145
110

130
100

120

110

90

80

500
250
125
85
65

1.1

28

Cr in

2,1 MN, NATIONAL R-100

520
260
130
85

7
14
21

C1

CERAMIC INSULATOR

1,000

µµf, 3.5 Mc

in ph,

A

R-1754

Plate

load (ohms)
C,

4 -LONG OR NATIONAL

140

3

3.3
2.5

4.1
3.1

higher.

For

70 -ohm

transmission line.

are for a Q of 12. for other values of Q, use
Values given are approximations. All components shown in Table
L,.
Q.
C.
Q.
is
higher
than 5,000 ohms, it is recommended that the
When the estimated plate load
and _
1

Q,,

-C,,

Q

L.

components be selected for a circuit Q between 20 and 30.

Table

1

Components for Pi- Coupled Final Amplifiers

To simplify design pro cedure, a pi- network chart,
compiled by M. Seybold,
W2RYI (reproduced by courtesy of R.C.A. Tube
Division, Harrison, N.J.) is shown in table 1.
This chart summarizes the calculations of figure 28 for various values of plate load.
Component Chart

for Pi- Networks

13 -11

Grid Bias

Radio-frequency amplifiers require some
form of grid bias for proper operation. Practically all r -f amplifiers operate in such a manner that plate current flows in the forni of
short pulses which have a duration of only a

fraction of an r -f cycle. To accomplish this

with a sinusoidal excitation voltage, the operating grid bias must be at least sufficient to
cut off the plate current. In very high efficiency Class C amplifiers the operating bias may
be many times the cutoff value. Cutoff bias,
it will be recalled, is that value of grid voltage which will reduce the plate current to
zero at the plate voltage employed. The method
for calculating it has been indicated previously. This theoretical value of cutoff will not
reduce the plate current completely to zero,
due to the variable-ft tendency or "knee"
which is characteristic of all tubes as the
cutoff point is approached.
Amplitude modulated Class C
amplifiers should be operated
with the grid bias adjusted to a value greater
than twice cutoff at the operating plate voltClass

C

Bias

Generation of

266

FOU

R-

F

THE

Energy

RADIO

FROLIC/RIVER

DRIVER

Figure 29
GRID -LEAK BIAS
The grid leak on an amplifier or multiplier
stage may also be used as the shunt feed
impedance to the grid of the tube when o
high value of grid leak (greater than perhaps
20,000 ohms) is used. When a lower value of
grid leak is to be employed, an r -f choke
should be used between the grid of the tube
and the grid leak to reduce r -f losses in the

grid leak resistance.

age. This procedure will insure that the tube
is operating at a bias greater than cutoff when
the plate voltage is doubled on positive modulation peaks. C -w telegraph and FM transmitters can be operated with bias as low as
cutoff, if only limited excitation is available
and moderate plate efficiency is satisfactory.
In a c -w transmitter, the bias supply or resistor should be adjusted to the point which
will allow normal grid current to flow for the
particular amount of grid driving r -f power
available. This form of adjustment will allow
more output from the under -excited r -f amplifier than when higher bias is used with corresponding lower values of grid current. In any
event, the operating bias should be set at as
low a value as will give satisfactory operation, since harmonic generation in a stage increases rapidly as the bias is increased.

resistor can be connected
Class
C amplifier to provide grid -leak bias. This resistor, R, in figure 29, is part of the d -c path
Grid -Leak Bias

A

in the grid circuit of a

circuit.
The r -f excitation applied to the grid circuit of the tube causes a pulsating direct current to flow through the bias supply lead, due
to the rectifying action of the grid, and any
current flowing through R, produces a voltage
drop across that resistor. The grid of the tube
is positive for a short duration of each r -f
cycle, and draws electrons from the filament
or cathode of the tube during that time. These
electrons complete the circuit through the d -c
grid return. The voltage drop across the resistance in the grid return provides a negain the grid

tive bias for the grid.
Grid-leak bias automatically adjusts itself
over fairly wide variations of r -f excitation.
The value of grid-leak resistance should be
such that normal values of grid current will
flow at the maximum available amount of r -f

Figure 30
COMBINATION GRID -LEAK AND
FIXED BIAS
Grid -leak bias often is used in conjunction
with a fixed minimum value of power supply
bias. This arrangement permits the operating
bias to be established by the excitation energy, but in the absence of excitation the electrode currents to the tube will be held to safe
values by the fixed- minimum power supply
bias. If a relatively low value of grid leak
is to be used, an r -f choke should be connected between the grid of the tube and the
grid leak as discussed in figure 29.

excitation. Grid -leak bias cannot be used for
grid -modulated or linear amplifiers in which
the average d -c grid current is constantly
varying with modulation.
Safety Bias

Grid -leak bias alone provides no

protection against e x c e s s i v e
plate current in case of failure of the source
of r -f grid excitation. A C- battery or C -bias
supply can be connected in series with the
grid leak, as shown in figure 30. This fixed
protective bias will protect the tube in the
event of failure of grid excitation. "Zero- bias"
tubes do not require this bias source in addition to the grid leak, since their plate current
will drop to a safe value when the excitation
is removed.

resistor can be connected in
series with the cathode or center- tapped filament lead of an amplifier to secure automatic bias. The plate current flows
through this resistor, then back to the cathode
or filament, and the voltage drop across the
resistor can be applied to the grid circuit by
connecting the grid bias lead to the grounded
or power supply end of the resistor R, as shown
Cathode Bias

A

in figure 31.

The grounded (B- minus) end of the cathode

resistor is negative relative to the cathode
by an amount equal to the voltage drop across
the resistor. The value of resistance must be
so chosen that the sum of the desired grid
and plate current flowing through the resistor

will bias the tube for proper operation.
This type of bias is used more extensively
in audio-frequency than in radio -frequency amplifiers. The voltage drop across the resistor

www.americanradiohistory.com

Protective Circuits

HANDBOOK

267

PROM
DRIVER

Figure
Figure

31

R

RF STAGE WITH CATHODE BIAS
Cathode bias sometimes is advantageous for
use it on r -f stage that operates with a relatively small amount of r -f excitation.

must be subtracted from the total plate supply
voltage when calculating the power input to
the amplifier, and this loss of plate voltage

in an r-f amplifier may be excessive. A Class
A audio amplifier is biased only to approximately one -half cutoff, whereas an r -f amplifier
may be biased to twice cutoff, or more, and
thus the plate supply voltage loss may be a
large percentage of the total available voltage
when using low or medium It tubes.
Oftentimes just enough cathode bias is employed in an r -f amplifier to act as safety bias
to protect the tubes in case of excitation failure, with the rest of the bias coming from a

grid leak.

Separate Bias
Supply

An external supply often is
used for grid bias, as shown in

figure 32. Battery bias gives
very good voltage regulation and is satisfactory for grid- modulated or linear amplifiers,
which operate at low grid current. In the case
of Class C amplifiers which operate with high
grid current, battery bias is not satisfactory.
This direct current has a charging effect on
the dry batteries; after a few months of service
the cells will become unstable, bloated, and

noisy.
A separate

a -c operated power supply is
commonly used for grid bias. The bleeder resistance across the output of the filter can be
made sufficiently low in value that the grid
current of the amplifier will not appreciably
change the amount of negative grid -bias voltage. Alternately, a voltage regulated grid -bias
supply can be used. This type of bias supply
is used in Class B audio and Class B r-f linear amplifier service where the voltage regulation in the C-bias supply is important. For
a Class C amplifier, regulation is not so important, and an economical design of components in the power supply, therefore, can be
utilized. In this case, the bias voltage must
be adjusted with normal grid current flowing,
as the grid current will raise the bias con-

-F

32

STAGE WITH BATTERY

BIAS

Battery bias is seldom used, due to deterioration of the cells by the reverse grid current.
However, it may be used in certain special
applications, or the fixed bias voltage may
be supplied by a bias power supply.

siderably when it is flowing through the bias supply bleeder resistance.

13 -12

Protective Circuits for
Tetrode Transmitting Tubes

The tetrode transmitting tube requires three
operating voltages: grid bias, screen voltage,
and plate voltage. The current requirements of
these three operating voltages are somewhat
interdependent, and a change in potential of
one voltage will affect the current drain of the
tetrode in respect to the other two voltages.
In particular, if the grid excitation voltage is
interrupted as by keying action, or if the plate
supply is momentarily interrupted, the resulting
voltage or current surges in the screen circuit
are apt to permanently damage the tube.

simple method of obtaining screen voltage is by
means of a dropping resistor from the high voltage plate supply, as shown
in figure 33. Since the current drawn by the
screen is a function of the exciting voltage
applied to the tetrode, the screen voltage will
rise to equal the plate voltage under conditions of no exciting voltage. If the control grid
is overdriven, on the other hand, the screen
current may become excessive. In either case,
damage to the screen and its associated components may result. In addition, fluctuations
in the plate loading of the tetrode stage will
cause changes in the screen current of the
tube. This will result in screen voltage fluctuations due to the inherently poor voltage
regulation of the screen series dropping resistor. These effects become dangerous to tube
life if the plate voltage is greater than the
screen voltage by a factor of 2 or so.

The Series Screen
Supply

www.americanradiohistory.com

A

268

Generation of

R

-F

THE

Energy

RADIO

RFC

r

NEGATIVE
OPERATING

Figure 33

8/AS CUTS
OFF

DROPPING- RESISTOR SCREEN SUPPLY

The Clomp Tube

CLAMP{

rUBE

4B
CLAMP
TUBE

Figure 34
CLAMP -TUBE SCREEN SUPPLY

A clamp tube may be added

to the series screen supply,
as shown in figure 34. The clamp tube is normally cut off by virtue of the d -c grid bias drop
developed across the grid resistor of the tetrode tube. When excitation is removed from
the tetrode, no bias appears across the grid
resistor, and the clamp tube conducts heavily,
dropping the screen voltage to a safe value.
When excitation is applied to the tetrode the
clamp tube is inoperative, and fluctuations of

the plate loading of the tetrode
allow the screen voltage to rise to
value. Because of this factor, the
does not offer complete protection
rode.

tube could
a damaging
clamp tube
to the tet-

A low voltage
may be used

screen supply
of the
series screen dropping resistor. This will protect the screen circuit from
excessive voltages when the other tetrode
operating parameters shift. However, the screen
can be easily damaged if plate or bias voltage is removed from the tetrode, as the screen
current will reach high values and the screen
dissipation will be exceeded. If the screen
supply is capable of providing slightly more
screen voltage than the tetrode requires for
proper operation, a series wattage -limiting resistor may be added to the circuit as shown
in figure 35. With this resistor in the circuit
it is possible to apply excitation to the tetrode tube with screen voltage present (but in
the absence of plate voltage) and still not damage the screen of the tube. The value of the
resistor should be chosen so that the product
of the voltage applied to the screen of the
tetrode times the screen current never exceeds
the maximum rated screen dissipation of the
tube.
The Separate
Screen Supply

instead

piing. The latter is a special form of inductive coupling. The choice of a coupling method
depends upon the purpose for which it is to
be used.

Capacitive coupling between an
amplifier or doubler circuit and a
preceding driver stage is shown
in figure 36. The coupling capacitor, C, isolates the d -c plate supply from the next grid
and provides a low impedance path for the r-f
energy between the tube being driven and the
driver tube. This method of coupling is simple
and economical for low power. amplifier or exciter stages, but has certain disadvantages,
particularly for high frequency stages. The
grid leads in an amplifier should be as short
as possible, but this is difficult to attain in
the physical arrangement of a high power amplifier with respect to a capacitively- coupled
driver stage.

Capacitive
Coupling

One significant disadvantage of capacitive coupling
is the difficulty of adjusting
the load on the driver stage.
Impedance adjustment can be accomplished
by tapping the coupling lead a part of the way
down on the plate coil of the tuned stage of
the driver circuit; but often when this is done
Disadvantages of
Capacitive
Coupling

SERIES RESISTOR
LOW VOLTAGE

SCREEN SUPPLY

13 -13

+B

Interstage Coupling

Energy is usually coupled from one circuit
of a transmitter into another either by capacitive coupling, inductive coupling, or link cou-

Figure 35
PROTECTIVE WATTAGE -LIMITING RESISTOR FOR USE WITH LOW- VOLTAGE
SCREEN SUPPLY
A

www.americanradiohistory.com

HANDBOOK

Figure 36
CAPACITIVE INTERSTAGE COUPLING

parasitic oscillation will take place in the
stage being driven.
One main disadvantage of capacitive coupling lies in the fact that the grid -to- filament
capacitance of the driven tube is placed directly across the driver tuned circuit. This
condition sometimes makes the r -f amplifier
difficult to neutralize, and the increased minimum circuit capacitance makes it difficult to
use a reasonable size coil in the v -h -f range.
Difficulties from this source can be partially
eliminated by using a center-tapped or split stator tank circuit in the plate of the driver
stage, and coupling capacitively to the opposite end from the plate. This method places
the plate -to- filament capacitance of the driver
across one -half of the tank and the grid -tofilament capacitance of the following stage
across the other half. This type of coupling is
a

shown in figure 37.
Capacitive coupling can be used to advantage in reducing the total number of tuned circuits in a transmitter so as to conserve space
and cost. It also can be used to advantage between stages for driving beam tetrode or pentode amplifier or doubler stages.
Inductive coupling (figure 38) results when two coils are electromagnetically coupled to one another. The degree of coupling is controlled by
varying the mutual inductance of the two coils,
which is accomplished by changing the spacing or the relationship between the axes of
the coils.

Inductive

Coupling

Interstage

Coupling

269

Figure 37
BALANCED CAPACITIVE COUPLING
Balanced capacitive coupling sometimes is
useful when it is desirable to use o relatively
large inductance in the interstage tank circuit, or where the exciting stage is neutralized as shown above.

Inductive coupling is used extensively for
coupling r -f amplifiers in radio receivers. However, the mechanical problems involved in adjusting the degree of coupling limit the usefulness of direct inductive coupling in transmitters. Either the primary or the secondary
or both coils may be tuned.
If the grid tuning capacitor of
figure 38 is removed and the
coupling increased to the maximum practicable

Unity Coupling

value by interwinding the turns of the two coils,
the circuit insofar as r.f. is concerned acts
like that of figure 36, in which one tank serves
both as plate tank for the driver and grid tank
for the driven stage. The inter-wound grid
winding serves simply to isolate the d-c plate
voltage of the driver from the grid of the driven
stage, and to provide a return for d -c grid current. This type of coupling, illustrated in figure 39, is commonly known as unity coupling.
Because of the high mutual inductance, both
primary and secondary are resonated by the
one tuning capacitor.
INTERWOUND

Figure 39

"UNITY" INDUCTIVE COUPLING

Figure 38
INDUCTIVE INTERSTAGE COUPLING

Due to the high value of coupling between
the two coils, one tuning capacitor tunes
both circuits. This arrangement often is usa
ful in coupling from a single -ended to a pushpull stage.

270

Generation of

R

-F

Energy

THE

LINK COUPLING

LINK COUPLING

AT

AT ..COLD. ENDS.
UPPER ENDS "MOT"

Figure 40
INTERSTAGE COUPLING BY MEANS
OF A

used since the two stages may be separated
by a considerable distance, since the amount
of a coupling between the two stages may he
easily varied, and since the capacitances of
the two stages may be isolated to permit use
of larger inductances in the v-h -f range.

special form of inductive
coupling which is widely employed in radio transmitter circuits is known
as link coupling. A low impedance r -f transmission line couples the two tuned circuits
together. Each end of the line is terminated
in one or more turns of wire, or links, wound
around the coils which are being coupled together. These links should be coupled to each
tuned circuit at the point of zero r -f potential,
or nodal point. A ground connection to one
side of the link usually is used to reduce harmonic coupling, or where capacitive coupling
between two circuits must be minimized. Coaxial line is commonly used to transfer energy
between the two coupling links, although Twin Lead may be used where harmonic attenuation
is not so important.
Typical link coupled circuits are shown in
figures 40 and 41. Some of the advantages of
link coupling are the following:
(1)
(2)

(3)

(4)
(5)

(6)

A

It eliminates coupling taps on tuned cir-

cuits.
It permits the use of series power supply
connections in both tuned grid and tuned
plate circuits, and thereby eliminates the
need of shunt -feed r -f chokes.
It allows considerable separation between
transmitter stages without appreciable
r-f losses or stray chassis currents.
It reduces capacitive coupling and thereby makes neutralization more easily attainable in r -f amplifiers.
It provides semi- automatic impedance
matching between plate and grid tuned
circuits, with the result that greater grid
drive can be obtained in comparison to
capacitive coupling.
It effectively reduces the coupling of harmonic energy.

COLO CENTER
ENDS "HOT*

Figure

"LINK"

PUSH -PULL

Link interstoge coupling is very commonly

Link Coupling

RADIO

41

LINK COUPLING

The link -coupling line and links can be
made of no. 18 push -back wire for coupling
between low -power stages. For coupling between higher powered stages the 150 -ohm
Twin -Lead transmission line is quite effective
and has very low loss. Coaxial transmission is
most satisfactory between high powered amplifier stages, and should always be used
where harmonic attenuation is important.

13 -14

Radio- Frequency
Chokes

Radio -frequency

chokes are connected in

circuits for the purpose of stopping the passage of r -f energy while still permitting a direct current or audio -frequency current to pass.
They consist of inductances wound with a
large number of turns, either in the form of a
solenoid, a series of solenoids, a single universal pie winding, or a series of pie windings. These inductors are designed to have as
much inductance and as little distributed or
shunt capacitance as possible. The unavoidable small amount of distributed capacitance
resonates the inductance, and this frequency
normally should be much lower than the frequency at which the transmitter or receiver
circuit is operating. R -f chokes for operation
on several bands must be designed carefully
so that the impedance of the choke will be extremely high (several hundred thousand ohms)
in each of the bands.
The direct current which flows through the
r -f choke largely determines the size of wire
to be used in the winding. The inductance of
r -f chokes for the v -h -f range is much less
than for chokes designed for broadcast and
ordinary short -wave operation. A very high
inductance r -f choke has more distributed capacitance than a smaller one, with the result

www.americanradiohistory.com

HANDBOOK

Shunt

+5G +11V

+SG

PARALLEL PLATE FEED

and

Series

Feed

271

+Nv

-BIAS

-BIAS

SERIES PLATE FEED

PARALLEL BIAS FEED

SERIES BIAS FEED

Figure 42

ILLUSTRATING PARALLEL AND
SERIES PLATE FEED
Parallel plate feed is desirable from a safety
standpoint since the tank circuit is at ground
potential with respect to d.c. However, a

Figure 43

ILLUSTRATING SERIES AND
PARALLEL BIAS FEED

high- impedance r-f choke is required, and
the r -t choke must be able to withstand the
peak r -f voltage output of the tube. Series

plate feed eliminates the requirement for a
high -performance r-f choke, but requires the
use of a relatively large value of by-pass

capacitance at the bottom end of the tank
circuit, as contrasted to the moderate value
of coupling capacitance which may be used
at the top of the tank circuit for parallel
plate feed.

that it will actually offer less impedance at
very high frequencies.

Another consideration, just as important as
the amount of d.c. the winding will carry, is
the r -f voltage which may be placed across
the choke without its breaking down. This is
a function of insulation, turn spacing, frequency, number and spacing of pies and other factors.
Some chokes which are designed to have a
high impedance over a very wide range of frequency are, in effect, really two chokes: a
u -h -f choke in series with a high -frequency
choke. A choke of this type is polarized; that
is, it is important that the correct end of the
combination choke be connected to the "hot"
side of the circuit.

Direct-current grid and plate
connections are made either by
series or parallel leed systems.
Simplified forms of each are shown in figures

Shunt and
Series Feed

42 and 43.

Series feed can be defined as that in which
the d -c connection is made to the grid or plate
circuits at a point of very low r-f potential.
Shunt feed always is made to a point of high
r -f voltage and always requires a high impedance r-f choke or a relatively high resistance
to prevent waste of r -f power.

Parallel and

13 -15

Push -Pull Tube Circuits
The comparative r -f power output from parallel or push -pull operated amplifiers is the same
if proper impedance matching is accomplished,
if sufficient grid excitation is available in
both cases, and if the frequency of measurement is considerably lower than the frequency
limit of the tubes.

Operating tubes in parallel has
some advantages in transmitters
designed for operation below 10
NIc., particularly when tetrode or pentode tubes
are to be used. Only one neutralizing capacitor
is required for parallel operation of triode
tubes, as against two for push -pull. Above
about 10 etc., depending upon the tube type,
parallel tube operation is not ordinarily recommended with triode tubes. However, parallel
operation of grounded -grid stages and stages
using low -C beam tetrodes often will give excellent results well into the v -h -f range.
Parallel
Operation

Push -Pull
Operation

The push -pull connection provides

well -balanced circuit insofar as
miscellaneous capacitances are
concerned; in addition, the circuit can be neutralized more completely, especially in high frequency amplifiers. The L/C ratio in a push pull amplifier can be made higher than in a
plate- neutralized parallel -tube operated amplifier. Push -pull amplifiers, when perfectly
balanced, have less second-harmonic output
than parallel or single -tube amplifiers, but in
practice undesired capacitive coupling and
circuit unbalance more or less offset the theoretical harmonic-reducing advantages of push pull r -f circuits.
a

CHAPTER FOURTEEN

R -F

Feedback

Comparatively high gain is required in single sideband equipment because the signal is
usually generated at levels of one watt or less.
To get from this level to a kilowatt requires
about 30 db of gain. High gain tetrodes may
be used to obtain this increase with a minimum
number of stages and circuits. Each stage contributes some distortion; therefore, it is good
practice to keep the number of stages to a
minimum. It is generally considered good practice to operate the low level amplifiers below
their maximum power capability in order to
confine most of the distortion to the last two
amplifier stages. R -f feedback can then be
utilized to reduce the distortion in the last
two stages. This type of feedback is no different from the common audio feedback used
in high fidelity sound systems. A sample of
the output waveform is applied to the amplifier input to correct the distortion developed
in the amplifier. The same advantages can be
obtained at radio frequencies that are obtained
at audio frequencies when feedback is used.

14 -1

R -F Feedback
Circuits

R -f feedback circuits have been developed
by the Collins Radio Co. for use with linear

amplifiers. Tests with large receiving and small
transmitting tubes showed that amplifiers using these tubes without feedback developed
signal -to- distortion ratios no better than 30 db
or so. Tests were run employing cathode follower circuits, such as shown in figure 1A.
Lower distortion was achieved, but at the cost
of low gain per stage. Since the voltage gain
through the tube is less than unity, all gain
has to be achieved by voltage step -up in the
tank circuits. This gain is limited by the dissipation of the tank coils, since the circuit
capacitance across the coils in a typical transmitter is quite high. In addition, the tuning
of such a stage is sharp because of the high
Q circuits.
The cathode follower performance of the
tube can be retained by moving the r -f ground

B
Br

Bi>5

`J

Figure 1
SIMILAR CATHODE FOLLOWER CIRCUITS HAVING DIFFERENT

272

www.americanradiohistory.com

R -F

GROUND POINTS.

R -F

Feedback Circuits

273

B

B*

BIAS

R -F

Tuning and loading are accomplished by Cr
and C,. C, and L, are tuned in unison to
establish the correct degree of feedback.

R F OUT

BIAS

=

B4

Figure 4
AMPLIFIER WITH FEEDBACK
AND IMPEDANCE MATCHING
OUTPUT NETWORK.

Figure 2
SINGLE STAGE AMPLIFIER WITH
R -F FEEDBACK CIRCUIT

B}

Figure 3
SINGLE STAGE FEEDBACK
AMPLIFIER WITH GROUND
RETURN POINT MODIFIED FOR

UNBALANCED INPUT AND
OUTPUT CONNECTIONS.

point of the circuit from the plate to the cathode as shown in figure 1B. Both ends of the
input circuit are at high r -f potential so inductive coupling to this type of amplifier is
necessary.

Inspection of figure 1B shows that by moving the top end of the input tank down on a
voltage divider tap across the plate tank circuit, the feedback can be reduced from 100%,
as in the case of the cathode follower circuit,
down to any desired value. A typical feedback
circuit is illustrated in figure 2. This circuit
is more practical than those of figure 1, since
the losses in the input tank are greatly reduced.
A feedback level of 12 db may be achieved
as a good compromise between distortion and
stage gain. The voltage developed across C=
will be three times the grid- cathode voltage.

Inductive coupling is required for this circuit, as shown in the illustration.
The circuit of figure 3 eliminates the need
for inductive coupling by moving the r -f
ground to the point common to both tank
circuits. The advantages of direct coupling between stages far outweigh the disadvantages of
having the r -f feedback voltage appear on the
cathode of the amplifier tube.
In order to match the amplifier to a load,
the circuit of figure 4 may be used. The ratio
of XL, to XC, determines the degree of feedback, so it is necessary to tune them in unison
when the frequency of operation is changed.
Tuning and loading functions are accomplished
by varying C2 and G. L5 may also be varied to
adjust the loading.
Feedback Around o
Two -Stage Amplifier

The maximum phase
shift obtainable over
two simple tuned cir-

cuits does not exceed 180 degrees, and feedback around a two stage amplifier is possible.
The basic circuit of a two stage feedback
amplifier is shown in figure 5. This circuit
is a conventional two -stage tetrode amplifier
except that r -f is fed back from the plate
circuit of the PA tube to the cathode of the
driver tube. This will reduce the distortion

1
E
Figure 5
BASIC CIRCUIT OF TWO -STAGE AMPLIFIER WITH R -F FEEDBACK
Feedback voltage is obtained from a voltage divider across the output circuit and
applied directly to the cathode of the first tube. The input tank circuit is thus
outside the feedback loop.

274

R -F

THE RADIO

Feedback

of both tubes as effectively as using individual
feedback loops around each stage, yet will
allow a higher level of overall gain. With
only two tuned circuits in the feedback loop,
it is possible to use 12 to 15 db of feedback
and still leave a wide margin for stability. It
is possible to reduce the distortion by nearly
as many db as are used in feedback. This circuit has two advantages that are lacking in the
single stage feedback amplifier. First, the filament of the output stage can now be operated
at r -f ground potential. Second, any conventional pi output network may be used.
R -f feedback will correct several types of
distortion. It will help correct distortion caused
by poor power supply regulation, too low grid
bias, and limiting on peaks when the plate
voltage swing becomes too high.
Neutralization

The purpose of neutralization of an r -f amplifier
stage is to balance out effects of the grid -plate capacitance coupling in
the amplifier. In a conventional amplifier using a tetrode tube, the effective input capacity
is given by:
ond

R -F

Feedback

Input Capacitance
where: Ci,,

C..
A

=

Cis

+ Cy.

(1

+A

cos e )

= tube input capacitance
= grid -plate capacitance
= voltage amplification from grid
to plate

e

= phase angle of load

In a typical unneutralized tetrode amplifier
having a stage gain of 33, the input capacitance of the tube with the plate circuit in
resonance is increased by 8 µµfd. due to the
unneutralized grid -plate capacitance. This is
unimportant in amplifiers where the gain (A )
remains constant but if the tube gain varies,
serious detuning and r -f phase shift may result.
A grid or screen modulated r -f amplifier is an
example of the case where the stage gain varies from a maximum down to zero. The gain
of a tetrode r -f amplifier operating below plate
current saturation varies with loading so that
if it drives a following stage into grid current
the loading increases and the gain falls off.
The input of the grid circuit is also affected
by the grid -plate capacitance, as shown in this
equation:

Input Resistance

-

27rf

X

C.N (

Asine

)

This resistance is in shunt with the grid
current loading, grid tank circuit losses, and
driving source impedance. When the plate cir-

cuit is inductive there is energy transferred
from the plate to the grid circuit (positive
feedback ) which will introduce negative resistance in the grid circuit. When this shunt
negative resistance across the grid circuit is
lower than the equivalent positive resistance
of the grid loading, circuit losses, and driving
source impedance, the amplifier will oscillate.
When the plate circuit is in resonance
( phase angle equal to zero) the input resistance due to the grid -plate capacitance becomes
infinite. As the plate circuit is tuned to the
capacitive side of resonance, the input resistance becomes positive and power is actually
transferred from the grid to the plate circuit.
This is the reason that the grid current in an
unneutralized tetrode r -f amplifier varies from
a low value with the plate circuit tuned on the
low frequency side of resonance to a high value
on the high frequency side of resonance The
grid current is proportional to the r -f voltage
on the grid which is varying under these conditions. In a tetrode class All amplifier, the
effect of grid -plate feedback can be observed
by placing a r -f voltmeter across the grid circuit and observing the voltage change as the
plate circuit is tuned through resonance.
If the amplifier is over -neutralized, the effects reverse so that with the plate circuit
tuned to the low frequency side of resonance
the grid voltage is high, and on the high frequency side of resonance, it is low.

Amplifier
Neutralization Check

useful "rule of
thumb" method of
checking neutralization of an amplifier stage (assuming that it
is nearly correct to start with) is to tune both
grid and plate circuits to resonance. Then, observing the r -f grid current, tune the plate circuit to the high frequency side of resonance.
If the grid current rises, more neutralization
capacitance is required. Conversely, if the grid
current decreases, less capacitance is needed.
This indication is very sensitive in a neutralized triode amplifier, and correct neutralization exists when the grid current peaks at the
point of plate current dip. In tetrode power
amplifiers this indication is less pronounced.
Sometimes in a supposedly neutralized tetrode
amplifier, there is practically no change in
grid voltage as the plate circuit is tuned
through resonance, and in some amplifiers it
is unchanged on one side of resonance and
drops slightly on the other side. Another observation sometimes made is a small dip in
the center of a broad peak of grid current.
These various effects are probably caused by

www.americanradiohistory.com

A

HANDBOOK

R -F

R

-F

Figure 7

NEUTRALIZED AMPLIFIER AND
INHERENT FEEDBACK CIRCUIT.
Neutralization

is achieved by varying
the capacity of Cn.

-

coupling from the plate to the grid circuit
through other paths which are not balanced
out by the particular neutralizing circuit used.
Figure 6 shows an r -f amplifier with negative feed of a One -Stage
back. The voltage developed
R -F Amplifier
across G due to the voltage
divider action of G and C,
is introduced in series with the voltage developed across the grid tank circuit and is in
phase- opposition to it. The feedback can be
made any value from zero to 100% by properly choosing the values of C:. and G.
For reasons stated previously, it is necessary
to neutralize this amplifier, and the relationship for neutralization is:
Feedback and

Neutralization

G

==

275

nur

Figure 6
SINGLE STAGE R -F AMPLIFIER
WITH FEEDBACK RATIO OF
DETERMINES
C C to C
C
STAGE NEUTRALIZATION

G

Feedback Circuits

GP

G.

It is often necessary to add capacitance from
plate to grid to satisfy this relationship
Figure 7 is identical to figure 6 except that
it is redrawn to show the feedback inherent in
this neutralization circuit more clearly. G and
C replace G and C., and the main plate tank
tuning capacitance is G. The circuit of figure
7 presents a problem in coupling to the grid
circuit. Inductive coupling is ideal, but the
extra tank circuits complicate the tuning of a
transmitter which uses several cascaded amplifiers with feedback around each one. The
grid could be coupled to a high source impedance such as a tetrode plate, but the driver
then cannot use feedback because this would
cause the source impedance to be low. A possible solution is to move the circuit ground
point from the cathode to the bottom end of

the grid tank circuit. The feedback voltage then
appears between the cathode and ground
( figure 8 ) . The input can be capacitively
coupled, and the plate of the amplifier can
be capacitively coupled to the next stage. Also,
cathode type transmitting tubes are available
that allow the heater to remain at ground po-

tential when r -f is impressed upon the cathode.
The output voltage available with capacity
coupling, of course, is less than the plate cathode r -f voltage developed by the amount
of feedback voltage across G.

14 -2

Feedback and

Neutralization of
Two -Stage

R -F

a

Amplifier

Feedback around two r -f stages has the advantage that more of the rube gain can be
realized and nearly as much distortion reduction can be obtained using 12 db around two
stages as is realized using 12 db around each
of two stages separately. Figure 9 shows a
basic circuit of a two stage feedback amplifier. Inductive output coupling is used, although a pi- network configuration will also
work well. The small feedback voltage required
is obtained from the voltage divider C. - G
and is applied to the cathode of the driver
tube. C. is only a few gcfd., so this feedback
voltage divider may be left fixed for a wide
frequency range. If the combined tube gain is
160, and 12 db of feedback is desired, the ratio
of G to C. is about 40 to 1. This ratio in
practice may be 400 µµtd. to 2.5 µµfd., for
example.
A complication is introduced into this simplified circuit by the cathode -grid capacitance
R

R-F.N

-i

Figure 8
UNBALANCED INPUT AND OUTPUT
CIRCUITS FOR SINGLE -STAGE
R -F AMPLIFIER WITH FEEDBACK

www.americanradiohistory.com

F

our

276

THE RADIO

Feedback

R -F

Figure 9
TWO -STAGE AMPLIFIER WITH FEEDBACK.
Included

is a

capacitor ,C) for neutralizing the cathode -grid capacity of the first tube.
by capacitor C and V; is neutralized by the correct ratio of C C-.

V.

is

neutralized

,

of the first tube which causes an undersired
coupling to the input grid circuit. It is necessary to neutralize out this capacitance coupling,
as illustrated in figure 9. The relationship for
neutralization is:

G

Cgt

G

G

The input circuit may be made unbalanced
by making C. five times the capacity of G.
This will tend to reduce the voltage across
the coil and to minimize the power dissipated
by the coil. For proper balance in this case,
G must be five times the grid -filament capacitance of the tube.
Except for tubes having extremely small
grid -plate capacitance, it is still necessary to
properly neutralize both tubes. If the ratio of
G to G is chosen to be equal to the ratio of
the grid -plate capacitance to the grid -filament
capacitance in the second tube (Vg), this tube
will be neutralized. Tubes such as a 4X -150A
have very low grid -plate capacitance and probably will not need to be neutralized when used
in the first (V.) stage. If neutralization is
necessary, capacitor G is added for this purpose and the proper value is given by the
following relationship:
CRO

G

_

Cet

G

G

C.

more feedback from the output stage to overcome.

Neutralizing the circuit of
figure 9 balances out coupling between the input
tank circuit and the output tank circuit, but it
does not remove all coupling from the plate
Tests For

Neutralization

circuit to the grid -cathode tube input. This
latter coupling is degenerative, so applying a
signal to the plate circuit will cause a signal
to appear between grid and cathode, even
though the stage is neutralized. A bench test
for neutralization is to apply a signal to the
plate of the tube and detect the presence of a
signal in the grid coil by inductive coupling
to it. No signal will be present when the stage
is neutralized. Of course, a signal could be inductively coupled to the input and neutralization accomplished by adjusting one branch of
the neutralizing circuit bridge (G for example) for minimum signal on the plate circuit.

Neutralizing the cathode -grid capacitance of
the first stage of figure 9 may be accomplished
by applying a signal to the cathode of the tube
and adjusting the bridge balance for minimum
signal on a detector inductively coupled to the
input coil.

Tuning the two -stage
feedback amplifier of
figure 9 is accomplished in an unconventional way because the
output circuit cannot be tuned for maximum
output signal. This is because the output circuit must be tuned so the feedback voltage
applied to the cathode is in -phase with the
input signal applied to the first grid. When
the feedback voltage is not in- phase, the resultTuning

o Two -Stage
Feedback Amplifier

If neither tube requires neutralization, the
bottom end of the interstage tank circuit may
be returned to r -f ground. The screen and
suppressor of the first tube should then be
grounded to keep the tank output capacitance directly across this interstage circuit and
to avoid common coupling between the feedback on the cathode and the interstage circuit.
A slight amount of degeneration occurs in the
first stage since the tube also acts as a grounded
grid amplifier with the screen as the grounded
grid. The p. of the screen is much lower than
that of the control grid so that this effect may
be unnoticed and would only require slightly

ant grid- cathode voltage increases as shown
in figure 10. When the output circuit is
properly tuned, the resultant grid -cathode voltage on the first tube will be at a minimum, and
the voltage on the interstage tuned circuit will
also be at a minimum.

www.americanradiohistory.com

HANDBOOK
1
VOLTAGE

VOLTAGE - GRID

TO

Neutralization

277

CATHODE

-

INPUT GRID
TO

GROUND

VOLTAGE - CATHODE TO GROUND
(PEEDBACM)

Figure 12
INTERSTAGE CIRCUIT WITH
SEPARATE NEUTRALIZING
AND FEEDBACK CIRCUITS.

(A.

A
B

Figure 10
VECTOR RELATIONSHIP OF
FEEDBACK VOLTAGE
Output Circuit Properly Tuned
Output Circuit Mis -Tuned

The two -stage amplifier may be tuned by
placing a r -f voltmeter across the interstage
tank circuit ( "hot" side to ground) and tuning
the input and interstage circuits for maximum
meter reading, and tuning the output circuit
for minimum meter reading. If the second tube
is driven into the grid current region, the grid
current meter may be used in place of the r -f
voltmeter. On high powered stages where operation is well into the Class AB region, the
plate current dip of the output tube indicates
correct output circuit tuning, as in the usual
amplifier.

Quite often low freq u e n c y parasitics
may be found in
the interstage circuit of the two -stage feedback
amplifier. Oscillation occurs in the first stage
due to low frequency feedback in the cathode
circuit. R -f chokes, coupling capacitors, and
bypass capacitors provide the low frequency
tank circuits. When the feedback and second
stage neutralizing circuits are combined, it is
necessary to use the configuration of figure 11.
This circuit has the advantage that only one
capacitor (G) is required from the plate of
the output tube, thus keeping the added capacitance across the output tank at a minimum.
Parasitic Oscillations in
the Feedback Amplifier

It is convenient, however, to separate these circuits so neutralization and feedback can be
adjusted independently. Also, it may be desirable to be able to switch the feedback out
of the circuit. For these reasons, the circuit
shown in figure 12 is often used. Switch S1
removes the feedback loop when it is closed.
A slight tendency for low frequency parasitic oscillations still exists with this circuit.
Li should have as little inductance as possible
without upsetting the feedback. If the value of
Li is too low, it cancels out part of the reactance of feedback capacitor G and causes
the feedback to increase at low values of radio
frequency. In some cases, a swamping resistor
may be necessary across L. The value of this
resistor should be high compared to the reactance of G to avoid phase -shift of the r -f
feedback.

Neutralization

14 -3

Procedure in
Feedback -Type Amplifiers
Experience with feedback amplifiers has
brought out several different methods of neutralizing. An important observation is that
when all three neutralizing adjustments are
correctly made the peaks and dips of various
tuning meters all coincide at the point of circuit resonance. For example, the coincident indications when the various tank circuits are
tuned through resonance with feedback operating are:
A -When the PA plate circuit is tuned

through resonance:

-PA plate current dip
2 -Power output peak
3 -PA r -f grid voltage dip
4 -PA grid current dip
1

+

BAS

Figure 11
INTERSTAGE CIRCUIT COMBINING

NEUTRALIZATION AND
FEEDBACK NETWORKS.

(Note: The PA grid current peaks
when feedback circuit is disabled
and the tube is heavily driven)

278

R -F

R-F

THE RADIO

Feedback

INc
Li o
O

°

cll
p FOUT

11(-

pi.

TY

gifIIN 1 c
C:7T
F

Tc.

ÌIr

ct0

HI,

your

RFC

1- 1-

o

T
1

l-

6+

BIAS

BIAS

Figure 13
TWO -STAGE AMPLIFIER WITH
B-When the PA grid circuit is tuned

through resonance:
Driver plate current dip
2 -PA r -f grid voltage peak
3 -PA grid current peak
4 -PA power output peak

1-

C -When the driver grid circuit is tuned

through resonance:

T

°

GF

FEED BACK

CIRCUIT.

2- Neutralize the grid -plate
tance of the driver stage
3- Neutralize the grid -plate

capaci-

capacitance of the power amplifier (PA)
stage
4 -Apply r -f feedback
Neutralize driver grid- cathode capacitance

5-

1-Driver r -f grid voltage peak
2-Driver plate current peak

These steps will be explained in more detail
in the following paragraphs:

3

Step 1. The removal of r -f feedback through
the feedback circuit must be complete. The
switch ( ) shown in the feedback circuit
( figure 13 ) is one satisfactory method. Since
C. is effectively across the PA plate tank circuit it is desirable to keep it across the circuit
when feedback is removed to avoid appreciable
detuning of the plate tank circuit. Another
method that can be used if properly done is
to ground the junction of C and C. Grounding this common point through a switch or
relay is not good enough because of common
coupling through the length of the grounding
lead. The grounding method shown in figure
14 is satisfactory.

-PA r -f grid current peak
-PA plate current peak
5 -PA power output peak
4

Four meters may be employed to measure
the most important of these parameters. The
meters should be arranged so that the following pairs of readings are displayed on meters
located close together for ease of observation
of coincident peaks and dips:
2

-PA plate current and power output
-PA r -f grid current and PA plate

3

-PA r -f grid

1

current

voltage and power output
Driver plate current and PA r-f
grid voltage

4-

The third pair listed above may not be
necessary if the PA plate current dip is pronounced. When this instrumentation is provided, the neutralizing procedure is as follows:

1- Remove the r-f feedback

FEEDBACK

Figure 14
SHORTING DEVICE.

Step 2. Plate power and excitation are applied.
The driver grid tank is resonated by tuning
for a peak in driver r -f grid voltage or driver
plate current. The power amplifier grid tank
circuit is then resonated and adjusted for a
dip in driver plate current. Driver neutralization is now adjusted until the PA r -f grid
voltage (or PA grid current) peaks at exactly
the point of driver plate current dip. A handy
rule for adjusting grid-plate neutralization of
a tube without feedback: with all circuits in
resonance, detune the plate circuit to the high
frequency side of resonance: If grid current
to next stage (or power output of the stage
under test) increases, more neutralizing capacitance is required and vice versa.

If the driver tube operates class A so that
a plate current dip cannot be observed, a dif-

www.americanradiohistory.com

HANDBOOK

Neutralization

279

Neutralization

The method of neutralization
employing a sensitive r -f detector inductively coupled to
a tank coil is difficult to apply in some cases
because of mechanical construction of the
equipment, or because of undesired coupling.
Another method for observing neutralization
can be used, which appears to be more accurate in actual practice. A sensitive r -f detector such as a receiver is loosely coupled to the
grid of the stage being neutralized, as shown
in figure 15. The coupling capacitance is of
the order of one or two µµfd. It must be small
enough to avoid upsetting the neutralization
when it is removed because the total grid ground capacitance is one leg of the neutralizing bridge. A signal generator is connected at
point S and the receiver at point R. If Coo is
not properly adjusted the S -meter on the receiver will either kick up or down as the grid
tank circuit is tuned through resonance. Go
may be adjusted for minimum deflection of the
S-meter as the grid circuit is tuned through
Techniques

T"
Figure 15
FEEDBACK NEUTRALIZING
CIRCUIT USING

AUXILIARY RECEIVER.

ferent neutralizing procedure is necessary This
will be discussed in a subsequent section.
Step 3. This is the same as step 2 except it
is applied to the power amplifier stage. Adjust the neutralization of this stage for a peak
in power output at the plate current dip.

Step 4. Reverse step
back.

1

and apply the r -f feed-

resonance.
Step 5. Apply plate power and an exciting signal to drive the amplifier to nearly full output. Adjust the feedback neutralization for a
peak in amplifier power output at the exact
point of minimum amplifier plate current.
Decrease the feedback neutralization capacitance if the power output rises when the tank
circuit is tuned to the high frequency side of
resonance.

The above sequence applies when the neutralizing adjustments are approximately correct to start with. If they are far off, some "cut and -try" adjustment may be necessary. Also,
the driver stage may break into oscillation if
the feedback neutralizing capacitance is not
near the correct setting.
It is assumed that a single tone test signal
for amplifier excitation during the
above steps, and that all tank circuits are at
resonance except the one being detuned to
make the observation. There is some interaction
between the driver neutralization and the feedback neutralization so if an appreciable change
is made in any adjustment the others should
be rechecked. It is important that the grid -plate
neutralization be accomplished first when using
is used

the above procedure, otherwise the feedback
neutralization will be off a little, since it partially compensates for that error.

The grid-plate capacitance of the tube is
then neutralized by connecting the signal generator to the plate of the tube and adjusting
Cti of figure 13 for minimum deflection again
as the grid tank is tuned through resonance.
The power amplifier stage is neutralized in
the same manner by connecting a receiver
loosely to the grid circuit, and attaching a
signal generator to the plate of the tube. The
r -f signal can be fed into the amplifier output
terminal if desired.
Some precautions are necessary when using
this neutralization method. First, some driver
tubes (the 6CL6, for example) have appreciably more effective input capacitance when
in operation and conducting plate current than
when in standby condition. This increase in
input capacitance may be as great as three or
four µµfd, and since this is part of the neutralizing bridge circuit it must be taken into
consideration. The result of this change in
input capacitance is that the neutralizing adjustment of such tubes must be made when
they are conducting normal plate current. Stray
coupling must be avoided, and it may prove
helpful to remove filament power from the
preceding stage or disable its input circuit in
some manner.
It should be noted that in each of the above
adjustments that minimum reaction on the
grid is desired, not minimum voltage. Some
residual voltage is inherent on the grid when
this neutralizing circuit is used.

CHAPTER FIFTEEN

Amplitude Modulation

If the output of a c -w transmitter is varied
in amplitude at an audio frequency rate instead of interrupted in accordance with code
characters, a tone will be heard on a receiver
tuned to the signal. If the audio signal consists of a band of audio frequencies comprising voice or music intelligence, then the
voice or music which is superimposed on the
radio frequency carrier will be heard on the

receiver.

When voice, music, video, or other intelligence is superimposed on a radio frequency
carrier by means of a corresponding variation
in the amplitude of the radio frequency output
of a transmitter, amplitude modulation is the
result. Telegraph keying of a c -w transmitter
is the simplest form of amplitude modulation,
while video modulation in a television transmitter represents a highly complex form. Systems for modulating the amplitude of a carrier
envelope in accordance with voice, music, or
similar types of complicated audio waveforms
are many and varied, and will be discussed
later on in this chapter.

15-1

Sidebands

Modulation is essentially a form of mixing
or combining already covered in a previous

chapter. To transmit voice at radio frequencies
by means of amplitude modulation, the voice

frequencies are mixed with a radio frequency
carrier so that the voice frequencies are converted to radio frequency sidebands. Though
it may be difficult to visualize, the amplitude
of the radio frequency carrier does not vary
during conventional amplitude modulation.
Even though the amplitude of radio frequency voltage representing the composite
signal ( resultant of the carrier and sidebands,
called the envelope) will vary from zero to
twice the unmodulated signal value during
full modulation, the amplitude of the carrier
component does not vary. Also, so long as
the amplitude of the modulating voltage does
not vary, the amplitude of the sidebands will
remain constant. For this to be apparent, however, it is necessary to measure the amplitude
of each component with a highly selective
filter. Otherwise, the measured power or voltage will be a resultant of two or more of the
components, and the amplitude of the resultant
will vary at the modulation rate.
If a carrier frequency of 5000 kc. is modulated by a pure tone of 1000 cycles, or 1 kc.,
two sidebands are formed: one at 5001 kc.
(the sum frequency) and one at 4999 kc. (the
difference frequency). The frequency of each
sideband is independent of the amplitude of
the modulating tone, or modulation percentage; the frequency of each sideband is determined only by the frequency of the modulating tone. This assumes, of course, that the
transmitter is not modulated in excess of its

linear capability.

280
www.americanradiohistory.com

Modulation
When the modulating signal consists of
multiple frequencies, as is the case with
voice or music modulation, two sidebands will
be formed by each modulating frequency (one
on each side of the carrier), and the radiated
signal will consist of a band of frequencies.
The band width, or channel taken up in the
frequency spectrum by a conventional double sideband amplitude-modulated signal, is equal
to twice the highest modulating frequency.
For example, if the highest modulating frequency is 5000 cycles, then the signal (assuming modulation of complex and varying
waveform) will occupy a band extending from
5000 cycles below the carrier to 5000 cycles
above the carrier.
Frequencies up to at least 2500 cycles, and
preferably 3500 cycles, are necessary for good
speech intelligibility. If a filter is incorporated in the audio system to cut out all frequencies above approximately 3000 cycles,
the band width of a radio- telephone signal can
be limited to 6 kc. without a significant loss
in intelligibility. However, if harmonic distortion is introduced subsequent to the filter, as
would happen in the case of an overloaded
modulator or overmodulation of the carrier,
new frequencies will be generated and the
signal will occupy a band wider than 6 kc.

Mechanics of
Modulation

15-2

fl

281

f
A

C.W. OR UNMODULATED CARRIER

SINE WAVE
AUDIO SIGNAL FROM MODULATOR

A 2

ItiÌ% Ì 1TjTI1 1ZIUIÌ
1111111111111111111111111
lA/2
_A /2
III 111

ÌI,

50

t

% MODULATED CARRIER

A
A

A

00%

MODULATED CARRIER

Figure
AMPLITUDE MODULATED WAVE
Top drawing (Al represents an unmodulated
carrier wave; (B) shows the audio output of
the modulator. Drawing (C) shows the audio
signal impressed on the carrier wave to the
extent of 50 per cent modulation; (D) shows
the carrier with 100 per cent amplitude modulation.
1

A c -w or unmodulated r-f carrier wave is
represented in figure 1A. An audio frequency
sine wave is represented by the curve of

figure

113.

When

the two are combined or

"mixed," the carrier is said to be amplitude
modulated, and a resultant similar to 1C or
is obtained. It should be noted that under
modulation, each half cycle of r -f voltage
differs slightly from the preceding one and
the following one; therefore at no time during
modulation is the r-f waveform a pure sine
wave. This is simply another way of saying
that during modulation, the transmitted r -f
energy no longer is confined to a single radio
frequency.
It will be noted that the average amplitude
of the peak r -f voltage, or modulation envelope, is the same with or without modulation.
This simply means that the modulation is
symmetrical (assuming a symmetrical modulating wave) and that for distortionless modulation the upward modulation is limited to a
value of twice the unmodulated carrier wave
amplitude because the amplitude cannot go
below zero on downward portions of the modulation cycle. Figure 1D illustrates the maxi1D

obtainable distortionless modulation with
a sine modulating wave, the r -f voltage at the
peak of the r -f cycle varying from zero to
twice the unmodulated value, and the r -f power
varying from zero to four times the unmodulated value ( the power varies as the square
of the voltage).
While the average r -f voltage of the modulated wave over a modulation cycle is the
mum

same as for the unmodulated carrier, the average power increases with modulation. If the
radio frequency power is integrated over the
audio cycle, it will be found with 100 per cent
sine wave modulation the average r-f power
has increased 50 per cent. This additional
power is represented by the sidebands, because as previously mentioned, the carrier
power does not vary under modulation. Thus,
when a 100 -watt carrier is modulated 100 per
cent by a sine wave, the total r -f power is 150
watts; 100 watts in the carrier and 25 watts
in each of the two sidebands.

www.americanradiohistory.com

282

Amplitude Modulation

THE

long as the relative proporLion of the various sidebands

Modulation
Percentage

RADIO

So

making up voice modulation is
maintained, the signal may be received and
detected without distortion. However, the
higher the average amplitude of the sidebands,
the greater the audio signal produced at the
receiver. For this reason it is desirable to
increase the modulation percentage, or degree
of modulation, to the point where maximum
peaks just hit 100 per cent. If the modulation
percentage is increased so that the peaks exceed this value, distortion is introduced, and
if carried very far, bad interference to signals
on nearby channels will result.
The amount by which a carrier
is being modulated may be expressed either as a modulation
factor, varying from zero to 1.0 at maximum
modulation, or as a percentage. The percentage of modulation is equal to 100 times the
modulation factor. Figure 2A shows a carrier
wave modulated by a sine -wave audio tone.
A picture such as this might be seen on the
screen of a cathode -ray oscilloscope with
sawtooth sweep on the horizontal plates and
the modulated carrier impressed on the vertical plates. The same carrier without modulation
would appear on the oscilloscope screen as

ECAR

Figure

2

GRAPHICAL DETERMINATION OF MODULATION PERCENTAGE
The procedure for determining modulation
percentage from the peak voltage points indicated is discussed in the text.

Modulation

Measurement

figure 2B.
The percentage of modulation of the positive peaks and the percentage of modulation
of the negative peaks can be determined separately from two oscilloscope pictures such
as shown.

The modulation factor of the positive peaks
may be determined by the formula:
Emax
M

=

-

Ecar

Ecar

The factor for negative peaks may be determined from this formula:
M

-

Ecar

-

Emin

Ecar
In the above two formulas Ern ax is the maximum carrier amplitude with modulation and
Ellin is the minimum amplitude; Ecar is the
steady -state amplitude of the carrier without modulation. Since the deflection of the
spot on a cathode -ray tube is linear with respect to voltage, the relative voltages of
these various amplitudes may be determined
by measuring the deflections, as viewed on
the screen, with a rule calibrated in inches
or centimeters. The percentage of modulation
of the carrier may be had by multiplying the
modulation factor thus obtained by 100. The
above procedure assumes that there is no

carrier shift, or change in average amplitude,
with modulation.
If the modulating voltage is symmetrical,
such as a sine wave, and modulation is accomplished without the introduction of distortion, then the percentage modulation will
be the same for both negative and positive
peaks. However, the distribution and phase
relationships of harmonics in voice and music
waveforms are such that the percentage modulation of the negative modulation peaks may
exceed the percentage modulation of the positive peaks, and vice versa. The percentage
modulation when referred to without regard
to polarity is an indication of the average of
the negative and positive peaks.
The modulation capability of a
transmitter is the maximum percentage to which that transmitter
may be modulated before spurious sidebands
are generated in the output or before the distortion of the modulating waveform becomes
objectionable. The highest modulation capability which any transmitter may have on the
negative peaks is 100 per cent. The maximum
permissible modulation of many transmitters
is less than 100 per cent, especially on positive peaks. The modulation capability of a
transmitter may be limited by tubes with inModulation

Capability

sufficient filament emission, by insufficient
excitation or grid bias to a plate- modulated
stage, too light loading of any type of amplifier carrying modulated r.f., insufficient power
output capability in the modulator, or too much
excitation to a grid-modulated stage or a
Class B linear amplifier. In any case, the
FCC regulations specify that no transmitter
be modulated in excess of its modulation
capability. Hence, it is desirable to make the
modulation capability of a transmitter as near
as possible to 100 per cent so that the carrier
power may be used most effectively.

www.americanradiohistory.com

HANDBOOK
Speech Waveform

Modulation Systems

The manner in which the
human voice is produced
by the vocal cords gives
rise to a certain dissymmetry in the waveform
of voice sounds when they are picked up by
a good -quality microphone. This is especially
pronounced in the male voice, and more so
on certain voiced sounds than on others. The
result of this dissymmetry in the waveform is
that the voltage peaks on one side of the
average value of the wave will be considerably greater, often two or three times as great,
as the voltage excursions on the other side
of the zero axis. The average value of voltage on both sides of the wave is, of course,
the same.
As a result of this dissymmetry in the male
voice waveform, there is an optimum polarity
of the modulating voltage that must be observed if maximum sideband energy is to be
obtained without negative peak clipping and
generation of "splatter" on adjacent channels.
A double -pole double -throw "phase reversing" switch in the input or output leads of any
transformer in the speech amplifier system will
permit poling the extended peaks in the direction of maximum modulation capability. The
optimum polarity may be determined easily by
listening on a selective receiver tuned to a
frequency 30 to 50 kc. removed from the desired signal and adjusting the phase reversing
switch to the position which gives the least
"splatter" when the transmitter is modulated
rather heavily. If desired, the switch then may
be replaced with permanent wiring, so long as
the microphone and speech system are not to
be changed.
A more conclusive illustration of the lopsidedness of a speech waveform may be obtained by observing the modulated waveform
of a radiotelephone transmitter on an oscilloscope. A portion of the carrier energy of the
transmitter should be coupled by means of a
link directly to the vertical plates of the
'scope, and the horizontal sweep should be a
sawtooth or similar wave occurring at a rate
of approximately 30 to 70 sweeps per second.
With the speech signal from the speech amplifier connected to the transmitter in one polarity it will be noticed that negative -peak
clipping -as indicated by bright "spots" in
the center of the 'scope pattern whenever the
carrier amplitude goes to zero -will occur at
a considerably lower level of average modulation than with the speech signal being fed to
the transmitter in the other polarity. When the
input signal to the transmitter is polarized in
such a manner that the "fingers" of the
speech wave extend in the direction of positive modulation these fingers usually will be
clipped in the plate circuit of the modulator
at an acceptable peak modulation level.
Dissymmetry

283

The use of the proper polarity of the incoming speech wave in modulating a transmitter
can afford an increase of approximately two
to one in the amount of speech audio power
which may be placed upon the carrier for an

amplitude-modulated transmitter for the same
amount of sideband splatter. More effective
methods for increasing the amount of audio
power on the carrier of an AM phone transmitter are discussed later in this chapter.

Because the same intelligibility is contained in each
of the sidebands associated
with a modulated carrier, it is not necessary
to transmit sidebands on both sides of the
carrier. Also, because the carrier is simply a
single radio frequency wave of unvarying amplitude, i t is no t necessary to transmit the
carrier if some means is provided for inserting
a locally generated carrier at the receiver.
When the carrier is suppressed but both
upper and lower sidebands are transmitted, it
is necessary to insert a locally generated
carrier at the receiver of exactly the same
frequency and phase as the carrier which was
suppressed. For this reason, suppressed carrier double -sideband systems have little
practical application.
When the carrier is suppressed and only the
upper or the lower sideband is transmitted, a
highly intelligible signal may be obtained at
the receiver even though the locally generated
carrier differs a few cycles from the frequency
of the carrier which was suppressed at the
transmitter. A communications system utilizing but one group of sidebands with carrier
suppressed is known as a single sideband
system. Such systems are widely used for
commercial point to point work, and are being
used to an increasing extent in amateur communication. The two chief advantages of the
system are: (1) an effective power gain of
about 9 db results from putting all the radiated power in intelligence carrying sideband
frequencies instead of mostly into radiated
carrier, and (2) elimination of the selective
fading and distortion that normally occurs in
a conventional double - sideband system when
the carrier fades and the sidebands do not, or
the sidebands fade differently.
Single- Sideband

Transmission

15 -3

Systems of Amplitude

Modulation
There are many different systems and methods for amplitude modulating a carrier, but
most may be grouped under three general classifications: (1) variable efficiency systems
in which the average input to the stage re-

www.americanradiohistory.com

284

THE

Amplitude Modulation

mains constant with and without modulation
and the variations in the efficiency of the
stage in accordance with the modulating signal accomplish the modulation; (2) constant
efficiency systems in which the input to the
stage is varied by an external source of modulating energy to accomplish the modulation;
and (3) so- called high -efficiency systems in
which circuit complexity is increased to obhigh plate circuit efficiency in the modulated
stage without the requirement of an external
high -level modulator. The various systems
under each classification have individual
characteristics which make certain ones best

suited to particular applications.

Since the average input
remains constant in a
stage employing variable
efficiency modulation, and since the average
power output of the stage increases with modulation, the additional average power output
from the stage with modulation must come from
the plate dissipation of the tubes in the stage.
Hence, for the best relation between tube cost
and power output the tubes employed should
have as high a plate dissipation rating per
Variable Efficiency
Modulation

dollar as possible.
The plate efficiency in such

an

ciency in certain types of amplifiers will be
as low as 60 per cent, the unmodulated efficiency in such amplifiers will be in the vicinity of 30 per cent.
Assuming a typical amplifier having a peak
efficiency of 70 per cent, the following figures give an idea of the operation of an idealized efficiency -modulated stage adjusted for
100 per cent sine -wave modulation. It should
be kept in mind that the plate voltage is constant at all times, even over the audio cycles.
100 watts
35 watts
35%

Input on 100% positive modulation
peak (plate current doubles)

Efficiency on

200 watts
70%

Output
tion peak

140

watts

0

watts

100% positive peak
on 100% positive modula-

Input on 100% negative peak
Efficiency on 100% negative peak
Output on 100% negative peak

0%
0 watts

100

watts

52.5 watts
52.5%

Systems of Efficiency

There are many systems of efficiency modulation, but they all
have the general limitation discussed in the
previous paragraph -so long as the carrier
amplitude is to remain constant with and
without modulation, the efficiency at carrier
level must be not greater than one -half the
peak modulation efficiency if the stage is to
be capable of 100 per cent modulation.
The classic example of efficiency modulation is the Class B linear r -f amplifier, to be
discussed below. The other three common
forms of efficiency modulation are control grid modulation, screen -grid modulation, and
suppressor -grid modulation. In each case,
including that of the Class B linear amplifier
note that the modulation, or the modulates
signal, is impressed on a control electrode
of the stage.
Modulation

amplifier is

when going from the unmodulated
condition to the peak of the modulation cycle.
Hence, the unmodulated efficiency of such an
amplifier must always be less than 45 per
cent, since the maximum peak efficiency obtainable in a conventional amplifier is in the
vicinity of 90 per cent. Since the peak effi-

doubled

Plate input without modulation
Output without modulation
Efficiency without modulation

Average input with 100%
modulation
Average output with 100% modulation (35 watts carrier plus 17.5
watts sideband)
Average efficiency with 100%
modulation

RADIO

The Class B

Linear Amplifier

This is the simplest practicable type amplifier for an

amplitude -modulated wave
or a single -sideband signal. The system possesses the disadvantage that excitation, grid
bias, and loading must be carefully controlled
to preserve the linearity of the stage. Also,
the grid circuit of the tube, in the usual application where grid current is drawn on peaks,
presents a widely varying value of load impedance to the source of excitation. Hence it
is necessary to include some sort of swamping
resistor to reduce the effect of grid- impedance variations with modulation. If such a
swamping resistance across the grid tank is
not included, or is too high in value, the positive modulation peaks of the incoming modulated signal will tend to be flattened with
resultant distortion of the wave being amplified.
The Class B linear amplifier has long been
used in broadcast transmitters, but recently
has received much more general usage in the
h -f range for two significant reasons: (a) the
Class B linear is an excellent way of increasing the power output of a single -sideband
transmitter, since the plate efficiency with
full signal will be in the vicinity of 70 per
cent, while with no modulation the input to
the stage drops to a relatively low value; and
(b) the Class B linear amplifier operates with
relatively low harmonic output since the grid
bias on the stage normally is slightly less

www.americanradiohistory.com

HANDBOOK

Class

than the value which will cut off plate current
to the stage in the absence of excitation.
Since s Class B linear amplifier is biased
to extended cutoff with no excitation ( the
grid bias at extended cutoff will be approximately equal to the plate voltage divided by
the amplification factor for a triode, and will
be approximately equal to the screen voltage
divided by the grid- screen mu factor for a
tetrode or pentode) the plate current will flow
essentially in 180 -degree pulses. Due to the
relatively large operating angle of plate current flow the theoretical peak plate efficiency
is limited to 78.5 per cent, with 65 to 70 per
cent representing a range of efficiency normally attainable, and the harmonic output

peak output of the r -f envelope should fall to
half the value obtained on positive modula-

will

be low.

The carrier power output from a Class B
linear amplifier of a normal 100 per cent modulated AM signal will be about one -half the
rated plate dissipation of the stage, with optimum operating conditions. The peak output
from a Class B linear, which represents the
maximum- signal output as a single -sideband
amplifier, or peak output with a 100 per cent
AM signal, will be about twice the plate dissipation of the tubes in the stage. Thus the
carrier -level input power to a Class B linear
should be about 1.5 times the rated plate dissipation of the stage.
The schematic circuit of a Class B linear
amplifier is the same as a conventional singleended or push -pull stage, whether triodes or
beam tetrodes are used. However, a swamping
resistor, as mentioned before, must be placed
across the grid tank of the stage if the operating conditions of the tube are such that
appreciable gridcurrent will be drawn on modulation peaks. Also, a fixed source of grid bias
must be provided for the stage. A regulated
grid -bias power supply is the usual source of

negative bias voltage.
With grid bias adjusted
to the correct value,
and with provision for
varying the excitation voltage to the stage
and the loading of the plate circuit, a fully
modulated signal is applied to the grid circuit
of the stage. Then with an oscilloscope coupled to the output of the stage, excitation and
loading are varied until the stage is drawing
the normal plate input and the output wave shape is a good replica of the input signal.
The adjustment procedure normally will reAdjustment of a Class
8 Linear Amplifier

quire a succession of approximations, until
the optimum set of adjustments is attained.
Then the modulation being applied to the input signal should be removed to check the
linearity. With modulation removed, in the
case of a 100 per cent AM signal, the input
to the stage should remain constant, and the

B

Linear Amplifier

285

tion peaks.
Class C
Grid Modulation
.

One widely used system of

efficiency

modulation for
communications
work
is
Class C control -grid bias modulation. The distortion is slightly higher than for a properly
operated Class B linear amplifier, but the efficiency is also higher, and the distortion can
be kept within tolerable limits for communications work.
Class C grid modulation requires high plate
voltage on the modulated stage, if maximum
output is desired. The plate voltage is normally run about 50 per cent higher than for
maximum output with plate modulation.
The driving power required for operation of
a grid -modulated amplifier under these conditions is somewhat more than is required for
operation at lower bias and plate voltage, but
the increased power output obtainable overbalances the additional excitation requirement. Actually, almost half as much excitation
is required as would be needed if the same
stage were to be operated as a Class C plate modulated amplifier. The resistor R across
the grid tank of the stage serves as swamping
to stabilize the r -f driving voltage. At least
50 per cent of the output of the driving stage
should be dissipated in this swamping resistor
under carrier conditions.
A comparatively small amount of audio power
will be required to modulate the amplifier stage
100 per cent. An audio amplifier having 20
watts output will be sufficient to modulate an
amplifier with one kilowatt input. Proportionately smaller amounts of audio will be required for lower powered stages. However, the
audio amplifier that is being used as the grid
modulator should, in any case, either employ
low plate resistance tubes such as 2A3's,
employ degenerative feedback from the output
stage to one of the preceding stages of the
speech amplifier, or be resistance loaded with
a resistor across the secondary of the modulation transformer. This provision of low drive
ing impedance in the grid modulator is to insure
good regulation in the audio driver for the grid
modulated stage. Good regulation of both the
audio and the r -f drivers of a grid -modulated
stage is quite important if distortion-free
modulation approaching 100 per cent is desired,
because the grid impedance of the modulated
stage varies widely over the audio cycle.
A practical circuit for obtaining grid -bias
modulation is shown in figure 3. The modulator and bias regulator tube have been combined in a single 6B4G tube.
The regulator -modulator tube operates as
a cathode - follower. The average d -c voltage

www.americanradiohistory.com

THE

Amplitude Modulation

286

R.F. AMPLIFIER

000

RFC

C

R

ANT

a

w.w.
#IOOA

The most satisfactory pro cedure for tuning a stage
for grid -bias modulation of
the Class C type is as
follows. The amplifier should first be neutralized, and any possible tendency toward parasitics under any condition of operation should
be eliminated. Then the antenna should be
coupled to the plate circuit, the grid bias
should be run up to the maximum available
value, and the plate voltage and excitation
should be applied. The grid bias voltage
should then be reduced until the amplifier
draws the approximate amount of plate current it is desired to run, and modulation corresponding to about 80 per cent is then applied.
If the plate current kicks up when modulation
is applied, the grid bias should be reduced;
if the plate meter kicks down, increase the

MIDGET CHOKE

.025
FROM
ETC

65J7

47K

R2
70 K

-

T

6UF.

it JO

5Y3GT

V.

325V.

sYQ0OO,
115

SMALL 60-80 MA.
V A

C

B C

Figure

TRANSFORMER

3

GRID -BIAS MODULATOR

per cent. If the antenna coupling is decreased
slightly from the condition just described, and
the excitation is increased to the point where
the amplifier draws the same input, carrier
efficiency of 50 per cent is obtainable with
tolerable distortion at 90 per cent modulation.
Tuning the
Grid -Bias
Modulated Stage

*5

25K IOW

AUDIO INPUT

RADIO

CIRCUIT

on the control grid is controlled by the 70, 000 ohm wire -wound potentiometer and this potentiometer adjusts the average grid bias on the

modulated stage. However, a -c signal voltage
is also impressed on the control -grid of the
tube and since the cathode follows this a -c
wave the incoming speech wave is superimposed on the average grid bias, thus effecting
grid -bias modulation of the r -f amplifier stage.
An audio voltage swing is required on the grid
of the 6B4G of approximately the same peak
value as will be required as bias -voltage
swing on the grid -bias modulated stage. This
voltage swing will normally be in the region
from 50 to 200 peak volts. Up to about 100
volts peak swing can be obtained from a 6SJ7
tube as a conventional speech amplifier stage.
The higher voltages may be obtained from a
tube such as a 6J5 through an audio transformer of 2:1 or 21/3:1 ratio.
With the normal amount of comparatively
tight antenna coupling to the modulated stage,
a non -modulated carrier efficiency of 40 per
cent can be obtained with substantially distortion -free modulation up to practically 100

grid bias.
When the amount of bias voltage has been
found (by adjusting the fine control, R2, on
the bias supply) where the plate meter remains constant with modulation, it is more
than probable that the stage will be drawing
either too much or too little input. The antenna coupling should then be either increased
or decreased (depending on whether the input was too little or too much, respectively)
until the input is more nearly the correct value.
The bias should then be readjusted until the
plate meter remains constant with modulation
as before. By slight jockeying back and forth
of antenna coupling and grid bias, a point can
be reached where the tubes are running at
rated plate dissipation, and where the plate
milliammeter on the modulated stage remains

substantially constant with modulation.
The linearity of the stage should then

be

checked by any of the conventional methods;
the trapezoidal pattern method employing a
cathode -ray oscilloscope is probably the most
satisfactory. The check with the trapezoidal
pattern will allow the determination of the
proper amount of gain to employ on the speech
amplifier. Too much audio power on the grid
of the modulated stage should not be used in
the tuning -up process, as the plate meter will

kick erratically and it will be impossible to
make a satisfactory adjustment.

Amplitude modulation may be
accomplished by varying the
screen -grid voltage in a Class
amplifier which employs a pentode, beam

Screen -Grid

Modulation
C

www.americanradiohistory.com

Screen Grid Modulation

H A N D B O O K

tetrode, or other type of screen -grid tube. The
modulation obtained in this way is not especially linear, but screen -grid modulation
does offer other advantages and the linearity
is quite adequate for communications work.
There are two significant and worthwhile
advantages of screen -grid modulation for communications work: (1) The excitation requirements for an amplifier which is to be modulated in the screen are not at all critical, and
good regulation of the excitation voltage is
not required. The normal rated grid- circuit
operating conditions specified for Class C
c -w operation are quite adequate for screen grid modulation. (2) The audio modulating
power requirements for screen -grid modulation
are relatively low.
A screen -grid modulated r -f amplifier operates as an efficiency -modulated amplifier, the
same as does a Class B linear amplifier and
a grid -modulated stage. Hence, plate circuit
loading is relatively critical as in any efficiency- modulated stage, and must be adjusted
to the correct value if normal power output
with full modulation capability is to be obtained. As in the case of any efficiency -modulated stage, the operating efficiency at the
peak of the modulation cycle will be between
70 and 80 per cent, with efficiency at the carrier level (if the stage is operating in the normal manner with full carrier) about half of the
peak- modulation value.
There are two main disadvantages of screen grid modulation, and several factors which
must be considered if satisfactory operation
of the screen -grid modulated stage is to be
obtained. The disadvantages are: (I) As mentioned before, the linearity of modulation with
respect to screen -grid voltage of such a stage
is satisfactory only for communications work,
unless carrier- rectified degenerative feed -back
is employed around the modulated stage to
straighten the linearity of modulation. (2) The
impedance of the screen grid to the modulating
signal is non -linear. This means that the modulating signal must be obtained from a source
of quite low impedance if audio distortion of
the signal appearing at the screen grid is to
be avoided.

Instead of being linear with respect to modulating voltage, as
is the plate circuit of a plate modulated Class C amplifier, the screen grid
presents approximately a square-law impedance to the modulating signal over the region
of signal excursion where the screen is positive with respect to ground. This non -linearity
may be explained in the following manner: At
the carrier level of a conventional screen modulated stage the plate -voltage swing of
the modulated tube is one -half the voltage
Screen -Grid
Impedance

287

swing at peak- modulation level. This condition
must exist in any type of conventional efficiency- modulated stage if 100 per cent positive modulation is to be attainable. Since the
plate -voltage swing is at half amplitude, and
since the screen voltage is at half its full modulation value, the screen current is relatively low. But at the positive modulation peak
the screen voltage is approximately doubled,
and the plate -voltage swing also is at twice
the carrier amplitude. Due to the increase in
plate -voltage swing with increasing screen
voltage, the screen current increases more than
linearly with increasing screen voltage.
In a test made on an amplifier with an 813
tube, the screen current at carrier level was
about 6 ma. with screen potential of 190 volts;
but under conditions which represented a positive modulation peak the screen current measured 25 ma. at a potential of 400 volts. Thus
instead of screen current doubling with twice
screen voltage as would be the case if the
screen presented a resistive impedance, the
screen current became about four times as
great with twice the screen voltage.
Another factor which must be considered
in the design of a screen -modulated stage, if
full modulation is to be obtained, is that the
power output of a screen -grid stage with zero
screen voltage is still relatively large. Hence,
if anything approaching full modulation on
negative peaks is to be obtained, the screen
potential must be made negative with respect
to ground on negative modulation peaks. In
the usual types of beam tetrode tubes the
screen potential must be 20 to 50 volts negative with respect to ground before cut -off of
output is obtained. This condition further complicates the problem of obtaining good linearity
in the audio modulating voltage for the screen modulated stage, since the screen voltage
must be driven negatively with respect to
ground over a portion of the cycle. Hence the
screen draws no current over a portion of the
modulating cycle, and over the major portion
of the cycle when the screen does draw current, it presents approximately a square -law
impedance.
Circuits for
ScreenGrid

Laboratory analysis of

a

large

number of circuits for accomModulation
plishing screen modulation has
led to the conclusion that the
audio modulating voltage must be obtained
from a low- impedance source if low- distortion modulation is to be obtained. Figure 4
shows a group of sketches of the modulation
envelope obtained with various types of modulators and also with insufficient antenna coupling. The result of this laboratory work led
to the conclusion that the cathode -follower
modulator of the basic circuit shown in figure

www.americanradiohistory.com

288

Amplitude Modulation

THE

RADIO

ENVELOPE OBTAINED WITH

INSUFFICIENT ANTENNA
COUPLING

+MOO.

-50

+5.0.

V.

APPROX.

O
Figure

4

SCREEN -MODULATION CIRCUITS
Three common screen modulation circuits are illustrated above. All three circuits
are capable of giving intelligible voice modulation although the waveform distortion
in the circuits of (A) and (B) is likely to be rather severe. The arrangement at (A)
is often called "clamp tube" screen modulation; by returning the grid leak on the
clomp tube to ground the circuit will give controlled- carrier screen modulation. This
circuit has the advantage that it is simple and is well suited to use in mobile transmitters. (B) is an arrangement using a transformer coupled modulator, and offers no
particular advantages. The arrangement at (C) is capable of giving good modulation
linearity due to the low impedance of the cathode-follower modulator. However, due
to the relatively low heater-cathode ratings on tubes suited for use as the modulator, a separate heater supply for the modulator tube normally is required. This limitation makes application of the circuit to the mobile transmitter a special problem,
since an isolated heater supply normally is not available. Shown at (D) as an assistance in the tuning of a screen -modulated transmitter (or any efficiency -modulated
transmitter for that matter) is the type of modulation envelope which results when
loading to the modulated stage is insufficient.

is capable of giving good -quality screen grid modulation, and in addition the circuit
provides convenient adjustments for the carrier level and the output level on negative
modulation peaks. This latter control, P2 in
figure 5, allows the amplifier to be adjusted
in such a manner that negative -peak clipping
cannot take place, yet the negative modulation
peaks may be adjusted to a level just above
that at which sideband splatter will occur.
5

The Cathode Follower Modulator

The cathode follower is
ideally suited for use as
the modulator for a screen-

grid stage since it acts as a relatively low impedance source of modulating voltage for
the screen -grid circuit. In addition the cathode follower modulator allows the supply voltage
both for the modulator and for the screen grid
of the modulated tube to be obtained from the
high -voltage supply for the plate of the screen grid tube or beam tetrode. In the usual case
the plate supply for the cathode follower, and
hence for the screen grid of the modulated
tube, may be taken from the bleeder on the
high- voltage power supply. A tap on the bleeder
may be used, or two resistors may be connected in series to make up the bleeder, with ap-

www.americanradiohistory.com

HANDBOOK

Modulation Systems

289

propriate values such that the voltage applied
to the plate of the cathode follower is appropriate for the tube to be modulated. It is important that a bypass capacitor be used from
the plate of the cathode - follower modulator
to ground.
The voltage applied to the plate of the
cathode follower should be about 100 volts
greater than the rated screen voltage for the
tetrode tube as a c -w Class C amplifier. Hence
the cathode -follower plate voltage should be
about 350 volts for an 815, 2E26, or 829B,
about 400 volts for an 807 or 4 -125A, about
500 volts for an 813, and about 600 volts for
a 4 -250A or a 4E27. Then potentiometer P1
in figure 5 should be adjusted until the carrier level screen voltage on the modulated stage
is about one -half the rated screen voltage
specified for the tube as a Class C c -w amplifier. The current taken by the screen of the
modulated tube under carrier conditions will
be about one - fourth the normal screen current
for c -w operation.
The only current taken by the cathode
follower itself will be that which will flow
through the 100,000 -ohm resistor between the
cathode of the 6L6 modulator and the negative supply. The current taken from the bleeder
on the high -voltage supply will be the carrier level screen current of the tube being modulated (which current passes of course through
the cathode follower) plus that current which
will pass through the 100,000 -ohm resistor.
The loading of the modulated stage should
be adjusted until the input to the tube is about
50 per cent greater than the rated plate dissipation of the tube or tubes in the stage. If the
carrier -level screen voltage value is correct
for linear modulation of the stage, the loading
will have to be somewhat greater than that
amount of loading which gives maximum output
from the stage. The stage may then be modulated by applying an audio signal to the grid
of the cathode -follower modulator, while observing the modulated envelope on an oscilloscope.
If good output is being obtained, and the
modulation envelope appears as shown in figure 4C, all is well, except that P2 in figure 5
should be adjusted until negative modulation
peaks, even with excessive modulating signal,
do not cause carrier cutoff with its attendant
sideband splatter. If the envelope appears as
at figure 4D, antenna coupling should be increased while the carrier level is backed down
by potentiometer PI in figure 5 until a set of

adjustments is obtained which will give a satisfactory modulation envelope as shown in
figure 4C.
Changing Bands

After a satisfactory set of adjustments has been obtained,

Figure 5
CATHODE -FOLLOWER
SCREEN -MODULATION CIRCUIT
A

detailed discussion of this circuit, which

also is represented in figure 4C, is given in
the accompanying text.

it is

not difficult to readjust the amplifier for
operation on different bands. Potentiometers
P1 (carrier level), and P2 ( negative peak level)
may be left fixed after a satisfactory adjustment, with the aid of the scope, has once been
found. Then when changing bands it is only
necessary to adjust excitation until the correct
value of grid current is obtained, and then to
adjust antenna coupling until correct plate
current is obtained. Note that the correct plate
current for an efficiency -modulated amplifier
is only slightly less than the out -of- resonance
plate current of the stage. Hence carrier -level
screen voltage must be low so that the out -ofresonance plate current will not be too high,
and relatively heavy antenna coupling must be
used so that the operating plate current will
be near the out -of- resonance value, and so that
the operating input will be slightly greater
than 1.5 times the rated plate dissipation of
the tube or tubes in the stage. Since the carrier
efficiency of the stage will be only 35 to 40
per cent, the tubes will be operating with plate
dissipation of approximately the rated value
without modulation.
Speech Clipping in

The maximum r -f output
of an efficiency -modulated stage is limited
by the maximum possible plate voltage swing
on positive modulation peaks. In the modula lation circuit of figure 5 the minimum output
is limited by the minimum voltage which the
screen will reach on a negative modulation
peak, as set by potentiometer P2 Hence
the
screen -grid- modulated stage, when using
the
modulator of figure 5, acts effectively as a
speech clipper, provided the modulating signal
amplitude is not too much more than that value
the Modulated Stage

www.americanradiohistory.com

290

THE

Amplitude Modulation

which will accomplish full modulation. With
correct adjustments of the operating conditions
of the stage it can be made to clip positive
and negative modulation peaks symmetrically.
However, the inherent peak clipping ability of
the stage should not be relied upon as a means
of obtaining a large amount of speech compression, since excessive audio distortion and
excessive screen current on the modulated
stage will result.
Characteristics of
Typical Screen
Modulated Stage

a

An

important character -

istic of the screen-modulated stage, when using

the cathode -follower modulator, is that excessive plate voltage on the
modulated stage is not required. In fact, full
output usually may be obtained with the larger
tubes at an operating plate voltage from one half to two- thirds the maximum rated plate
voltage for c -w operation. This desirable condition is the natural result of using a low impedance source of modulating signal for
the stage.
As an example of a typical screen -modulated stage, full output of 75 watts of carrier
may be obtained from an 813 tube operating
with a plate potential of only 1250 volts. No
increase in output from the 813 may be obtained by increasing the plate voltage, since
the tube may be operated with full rated plate
dissipation of 125 watts, with normal plate
efficiency for a screen -modulated stage, 37.5
per cent, at the 1250-volt potential.
The operating conditions of a screen -modulated 813 stage are as follows:

Plate voltage-1250 volts
Plate current -160 ma.
Plate input -200 watts
Grid current -11 ma.
Grid bias -I 10 volts
Carrier screen voltage -190 volts
Carrier screen current -6 ma.
Power output -approx. 75 watts

With full 100 per cent modulation the plate
current decreases about 2 ma. and the screen

current increases about 1 ma.; hence plate,
screen, and grid current remain essentially
constant with modulation. Referring to figure
5, which was the circuit used as modulator
for the 813, (El) measured plus 155 volts, (E2)
measured -50 volts, (E3) measured plus 190
volts, (Et) measured plus 500 volts, and the
r.m.s. swing at (E5) for full modulation measured 210 volts, which represents a peak swing
of about 296 volts. Due to the high positive
voltage, and the large audio swing, on the
cathode of the 6L6 (triode connected) modulator tube, it is important that the heater of
of this tube be fed from a separate filament

RADIO

transformer or filament winding. Note also that
the operating plate -to- cathode voltage on the
6L6 modulator tube does not exceed the 360 volt rating of the tube, since the operating
potential of the cathode is considerably above
ground potential.

Still another form of efficiency modulation may be
obtained by applying the
audio modulating signal to the suppressor grid
of a pentode Class C r -f amplifier. Basically,
suppressor -grid modulation operates in the
same general manner as other forms of efficiency modulation; carrier plate circuit efficiency is about 35 per cent, and antenna coupling must be rather tight. However, suppressor grid modulation has one sizeable disadvantage,
in addition to the fact that pentode tubes are
not nearly so widely used as beam tetrodes
which of course do not have the suppressor
element. This disadvantage is that the screen grid current to a suppressor -grid modulated
amplifier is rather high. The high screen current is a natural consequence of the rather high
negative bias on the suppressor grid, which
reduces the plate- voltage swing and plate current with a resulting increase in the screen
current.
In tuning a suppressor -grid modulated amplifier, the grid bias, grid current, screen voltage, and plate voltage are about the same as
for Class C c -w operation of the stage. But
the suppressor grid is biased negatively to a
value which reduces the plate- circuit efficiency to about one -half the maximum obtainable
from the particular amplifier, with antenna
coupling adjusted until the plate input is about
1.5 times the rated plate dissipation of the
stage. It is important that the input to the
screen grid be measured to make sure that the
rated screen dissipation of the tube is not
being exceeded. Then the audio signal is applied to the suppressor grid. In the normal
application the audio voltage swing on the
suppressor will be somewhat greater than the
negative bias on the element. Hence suppressor -grid current will flow on modulation
peaks, so that the source of audio signal voltage must have good regulation. Tubes suitable
for suppressor -grid modulation are: 2E22,
837, 4E27/8001, 5 -125, 804 and 803. A typical suppressor -grid modulated amplifier is
illustrated in figure 6.
Suppressor -Grid
Modulation

15 -4

Input Modulation
Systems

Constant efficiency variable -input modulation systems operate by virtue of the addition

www.americanradiohistory.com

HANDBOOK

Plate Modulation
CARRIER
OUTPUT

4E27

'33w

R.F INPUT

-

IG=

AIA

ISG'
44 M

-130

6J5

V.

2.1 STEPUP

IP=)OMA.

+1500

V.

PEAK SWING FOR FULL
MODULATION = 210 V.

A.F INPUT

+300 V

-210

Figure

V.

6

AMPLIFIER WITH SUPPRESSOR -GRID
MODULATION
Recommended operating conditions for linear suppressor-grid modulation of a 4E27/
2578/8001 stage are given on the drawing.

of external power to the modulated stage to
effect the modulation. There are two general
classifications that come under this heading;
those systems in which the additional power
is supplied as audio frequency energy from a
modulator, usually called plate modulation
systems, and those systems in which the additional power to effect modulation is supplied
as direct current from the plate supply.
Under the former classification comes Heising modulation (probably the oldest type of
modulation to be applied to a continuous carrier), Class B plate modulation, and series
modulation. These types of plate modulation
are by far the easiest to get into operation,
and they give a very good ratio of power input
to the modulated stage to power output; 65 to
80 per cent efficiency is the general rule. It
is for these two important reasons that these
modulation systems, particularly Class B plate
modulation, are at present the most popular
for communications work.
Modulation systems coming under the second classification are of comparatively recent
development but have been widely applied to
broadcast work. There are quite a few systems
in this class. Two of the more widely used
are the Doherty linear amplifier, and the Ter man- Woodyard high- efficiency grid- modulated
amplifier. Both systems operate by virtue of
a carrier amplifier and a peak amplifier connected together by electrical quarter -wave
lines. They will be described later in this
section.
Plate Modulation

Plate modulation is the application of the audio power

291

to the plate circuit of an r -f amplifier. The r -f
amplifier must be operated Class C for this

type of modulation in order to obtain a radio frequency output which changes in exact accordance with the variation in plate voltage.
The r -f ampli fier is 100 per cent modulated
when the peak a -c voltage from the modulator
is equal to the d.c. voltage applied to the r -f
tube. The positive peaks of audio voltage increase the instantaneous plate voltage on the
r -f tube to twice the .1c value, and the negative peaks reduce the voltage to zero.
The instantaneous plate current to the r -f
stage also varies in accordance with the modulating voltage. The peak alternating current
in the output of a modulator must be equal to
the d -c plate current of the Class C r -f stage
at the point of 100 per cent modulation. This
combination of change in audio voltage and
current can be most easily referred to in terms
of audio power in watts.
In a sinusoidally modulated wave, the antenna current increases approximately 22 per
cent for 100 per cent modulation with a pure
tone input; an r -f meter in the antenna circuit
indicates this increase in antenna current.
The average power of the r -f wave increases
50 per cent for 100 per cent modulation, the
efficiency remaining constant.
This indicates that in a plate- modulated
radiotelephone transmitter, the audio- frequency
channel must supply this additional 50 per
cent increase in average power for sine -wave
modulation. If the power input to the modulated stage is 100 watts, for example, the
average power will increase to 150 watts at
100 per cent modulation, and this additional
50 watts of power must be supplied by the
modulator when plate modulation is used. The
actual antenna power is a constant percentage
of the total value of input power.
One of the advantages of plate (or power)
modulation is the ease with which proper adjustments can be made in the transmitter. Also.
there is less plate loss in the r -f amplifier for
a given value of carrier power than with other
forms of modulation because the plate efficiency is higher.
By properly matching the plate impedance
of the r -f tube to the output of the modulator,
the ratio of voltage and current swing to d -c
voltage and current is automatically obtained.
The modulator should have a peak voltage
output equal to the average d -c plate voltage
on the modulated stage. The modulator should
also have a peak power output equal to the
d -c plate input power to the modulated stage.

The average power output of the modulator will
depend upon the type of waveform. If the amplifier is being Heising modulated by a Class
A stage, the modulator must have an average

www.americanradiohistory.com

292

CLASS

MODULATED CLASS C
R.

RADIO

THE

Amplitude Modulation
C

AMPLIFIER

f. AMPLIFIER

CLASS

IS

MODULATOR

+9

Figure

7

HEISING PLATE MODULATION
This type of modulation was the first form
of plate modulation. It is sometimes known
as "constant current" modulation. Because
of the effective 1:1 ratio of the coupling
choke, it is impossible to obtain 100 per cent
modulation unless the plate voltage to the
modulated stage is dropped slightly by resistor R. The capacitor C merely byp
the audio around R, so that the full a-f output voltage of the modulator is impressed
on the Class C stage.

power output capability of one -half the input
to the Class C stage. If the modulator is a
Class B audio amplifier, the average power
required of it may vary from one -quarter to more
than one -half the Class C input depending
upon the waveform. However, the peak power
output of any modulator must be equal to the
Class C input to be modulated.

Heising modulation is the oldest
system of plate modulation, and
usually consists of a Class A
audio amplifier coupled to the r -f amplifier by
means of a modulation choke coil, as shown
in figure 7.
The d.c. plate voltage and plate current in
the r-f amplifier must be adjusted to a value
which will cause the plate impedance to match
Heising
Modulation

the output of the modulator, since the modulation choke gives a 1 -to -1 coupling ratio. A
series resistor, by- passed for audio frequencies by means of a capacitor, must be connected in series with the plate of the r -f amplifier
to obtain modulation up to 100 per cent. The
peak output voltage of a Class A amplifier
does not reach a value equal to the d -c voltage
applied to the amplifier and, consequently,
the d -c plate voltage impressed across the
r -f tube must be reduced to a value equal to

MOD.

Figure

+5

R F.

.13

8

PLATE MODULATION
This type of modulation is the most flexible
in that the loading adjustment can be made
in a short period of time and without elaborate test equipment after a change in operating frequency of the Class C amplifier has
CLASS

B

been made.

if

available a -c peak voltage
100% modulation is to be obtained.
A higher degree of distortion can be tolerthe maximum

ated in low -power emergency phone transmitters
which use a pentode modulator tube, and the
series resistor and by -pass capacitor are
usually omitted in such transmitters.

High -level Class B plate
modulation is the least expensive method of plate
modulation. Figure 8 shows a conventional
Class B plate -modulated Class C amplifier.
The statement that the modulator output
power must be one -half the Class C input for
100 per cent modulation is correct only if the
waveform of the modulating power is a sine
wove. Where the modulator waveform is unclipped speech, the average modulator power
for 100 per cent modulation is considerably
less than one -half the Class C input.
Class B
Plata Modulation

It has been determined experimentally that the ratio
of peak to average power
in a speech waveform is approximately 4 to 1
as contrasted to a ratio of 2 to 1 in a sine
wave. This is due to the high harmonic content of such a waveform, and to the fact that
Power Relations in
Speech Waveforms

www.americanradiohistory.com

HANDBOOK

Plate Modulation

this high harmonic content manifests itself by
making the wave unsymmetrical and causing
sharp peaks or "fingers" of high energy content to appear. Thus for unclipped speech, the
average modulator plate current, plate dissipation, and power output are approximately
one -half the sine wave values for a given peak
output power.
Both peak power and average power are
necessarily associated with waveform. Peak
power is just what the name implies; the power
at the peak of a wave. Peak power, although
of the utmost importance in modulation, is of
no great significance in a -c power work, except insofar as the average power may be determined from the peak value of a known wave
form.

There is no time element implied in the
definition of peak power; peak power may be
instantaneous -and for this reason average
power, which is definitely associated with
time, is the important factor in plate dissipation. It is possible that the peak power of a
given waveform be several times the average
value; for a sine wave, the peak power is twice
the average value, and for unclipped speech
the peak power is approximately four times
the average value. For 100 per cent modulation, the peak (instantaneous) audio power
must equal the Class C input, although the
average power for this value of peak varies
widely depending upon the modulator waveform, being greater than 50 per cent for speech
that has been clipped and filtered, 50 per cent
for a sine wave, and about 25 per cent for typical unclipped speech tones.
Modulation
Transformer

Calculations

The

modulation

transformer is

a device for matching the load

impedance of the Class C amplifier to the recommended load
impedance of the Class B modulator tubes.
Modulation transformers intended for communications work are usually designed to
carry the Class C plate current through their
secondary windings, as shown in figure 8.
The manufacturer's ratings should be consulted to insure that the d-c plate current
passed through the secondary winding does
not exceed the maximum rating.
A detailed discussion of the method of
making modulation transformer calculations
has been given in Chapter Six. However, to
emphasize the method of making the calculation, an additional example will be given.
Suppose we take the case of a Class C amplifier operating at a plate voltage of 2000
with 225 ma. of plate current. This amplifier
would present a load resistance of 2000 divided by 0.225 amperes or 8888 ohms. The plate
power input would be 2000 times 0.225 or 450
watts. By reference to Chapter Six we see that

293

a pair of 811 tubes operating at 1500 plate
volts will deliver 225 watts of audio output.
The plate -to -plate load resistance for these

tubes under the specified operating conditions
is 18,000 ohms. Hence our problem is to match
the Class C amplifier load resistance of 8888
ohms to the 18,000 -ohm load resistance required by the modulator tubes.
A 200 -to -300 watt modulation transformer
will be required for the job. If the taps on the
transformer are given in terms of impedances
it will only be necessary to connect the secondary for 8888 ohms (or a value approximately
equal to this such as 9000 ohms) and the primary for 18,000 ohms. If it is necessary to
determine the proper turns ratio required of the
transformer it can be determined in the following manner. The square root of the impedance
ratio is equal to the turns ratio, hence:
8888
18000

=

V 0.494

=

0.703

The transformer must have a turns ratio of
approximately 1- to -0.7 step down, total primary to total secondary. The greater number
of turns always goes with the higher impedance, and vice versa.
Plate- andScreen
Modulation

only the plate of a
screen -grid tube is modulated, it is impossible to obtain high -percentage linear modulation under
ordinary conditions. The plate current of such
a stage is not linear with plate voltage. However, if the screen is modulated simultaneously
with the plate, the instantaneous screen voltage drops in proportion to the drop in the plate
voltage, and linear modulation can then be obtained. Four satisfactory circuits for accomplishing combined plate and screen modulaWhen

tion are shown in figure 9.
The screen r -f by -pass capacitor C2 should
not have a greater value than 0.005 µfd., preferably not larger than 0.001 tad. It should be
large enough to bypass effectively all r -f voltage without short- circuiting high- frequency
audio voltages. The plate by -pass capacitor
can be of any value from 0.002 µfd. to 0.005
µfd. The screen -dropping resistor, 111. should
reduce the applied high voltage to the value
specified for operating the particular tube in
the circuit. Capacitor C1 is seldom required
yet some tubes may require this capacitor in
order to keep C2 from attenuating the high frequencies. Different values between .0002 and
.002 µfd. should be tried for best results.
Figure 9C shows another method which uses
a third winding on the modulation transformer,
through which the screen -grid is connected to

www.americanradiohistory.com

294

RADIO

THE

Amplitude Modulation

E

3

3

B+ S.G.

Figure

B+

9

PLATE MODULATION OF A BEAM TETRODE OR SCREEN -GRID TUBE
These alternative arrangements for plate modulation of tetrodes or pentodes are discussed in detail in the text. The arrangements shown at (B) or (D) are recommended
for most applications.

a low- voltage power supply. The ratio of turns
between the two output windings depends upon
the type of screen -grid tube which is being
modulated. Normally it will be such that the
screen voltage is being modulated 60 per cent
when the plate voltage is receiving 100 per

cent modulation.
If the screen voltage is derived from a dropping resistor ( not a divider) that is bypassed
for r.f. but not a.f., it is possible to secure
quite good modulation by applying modulation
only to the plate. Under these conditions, the
screen tends to modulate itself, the screen
voltage varying over the audio cycle as a result of the screen impedance increasing with
plate voltage, and decreasing with a decrease

in plate voltage. This circuit arrangement is
illustrated in figure 9B.
A similar application of this principle is
shown in figure 9D. In this case the screen
voltage is fed directly from a low- voltage supply of the proper potential through a choke L.
A conventional filter choke having an inductance from 10 to 20 henries will be satisfactory for L.
To afford protection of the tube when plate
voltage is not applied but screen voltage is
supplied from the exciter power supply, when
using the arrangement of figure 9D, a resistor
of 3000 to 10,000 ohms can be connected in
series with the choke L. In this case the screen
supply voltage should be at least 1%Z times as

www.americanradiohistory.com

HANDBOOK

Cathode Modulation

much as is required tor actual screen

voltage,

and the value of resistor is chosen such that
with normal screen current the drop through
the resistor and choke will be such that normal screen voltage will be applied to the tube.
When the plate voltage is removed the screen
current will increase greatly and the drop
through resistor R will increase to such a
value that the screen voltage will be lowered
to the point where the screen dissipation on
the tube will not be exceeded. However, the
supply voltage and value of resistor R must
be chosen carefully so that the maximum rated
screen dissipation cannot be exceeded. The
maximum possible screen dissipation using
this arrangement is equal to: W = E' /4R where
E is the screen supply voltage and R is the
combined resistance of the resistor in figure
9D and the d -c resistance of the choke L. It
is wise, when using this arrangement to check,
using the above formula, to see that the value
of W' obtained is less than the maximum rated
screen dissipation of the tube or tubes used
in the modulated stage. This same system can
of course also be used in figuring the screen
supply circuit of a pentode or tetrode amplifier stage where modulation is not to be

applied.
The modulation transformer for plate -andscreen- modulation, when utilizing a dropping
resistor as shown in figure 9A, is similar to
the type of transformer used for any plate
modulated phone. The combined screen and
plate current is divided into the plate voltage
in order to obtain the Class C amplifier load
impedance. The peak audio power required to
obtain 100 per cent modulation is equal to the
d-c power input to the screen, screen resistor,
and plate of the modulated r -f stage.
15 -5

Cathode Modulation

Cathode modulation offers a workable compromise between the good plate efficiency but
expensive modulator of high -level plate modulation, and the poor plate efficiency but inexpensive modulator of grid modulation. Cathode
modulation consists essentially of an admixture of the two.
The efficiency of the average well- designed
plate -modulated transmitter is in the vicinity
of 75 to 80 per cent, with a compromise perhaps at 77.5 per cent. On the other hand, the
efficiency of a good grid-modulated transmitter
may run from 28 to maybe 40 per cent, with
the average falling at about 34 per cent. Now
since cathode modulation consists of simultaneous grid and plate modulation, in phase
with each other, we can theoretically obtain
any efficiency from about 34 to 77.5 per cent
from our cathode -modulated stage, depending

295

upon the relative percentages of grid and
plate modulation.
Since the system is a compromise between
the two fundamental modulation arrangements,
a value of efficiency approximately half way
between the two would seem to be the best
compromise. Experience has proved this to be
the case. A compromise efficiency of about
56.5 per cent, roughly half way between the
two limits, has proved to be optimum. Calculation has shown that this value of efficiency can be obtained from a cathode -modulated amplifier when the audio- frequency modulating power is approximately 20 per cent of
the d-c input to the cathode -modulated stage.

Series cathode modulation is
ideally suited as an economiModulator
cal modulating arrangement
for a high -power triode c -w
transmitter. The modulator can be constructed
quite compactly and for a minimum component
cost since no power supply is required for it.
When it is desired to change over from c -w to
'phone, it is only necessary to cut the series
modulator into the cathode return circuit of the
c -w amplifier stage. The plate voltage for the
modulator tubes and for the speech amplifier
is taken from the cathode voltage drop of the
modulated stage across the modulator unit.
Figure 10 shows the circuit of such a modulator, designed to cathode modulate a Class C
amplifier using push -pull 810 tubes, running
at a supply voltage of 2500, and with a plate
input of 660 watts. The modulated stage runs
at about 50% efficiency, giving a power output
of nearly 350 watts, fully modulated. The voltage drop across the cathode modulator is 400
volts, allowing a net plate to cathode voltage
of 2100 volts on the final amplifier. The plate
current of the 810's should be about 330 ma.,
and the grid current should be approximately
40 ma., making the total cathode current of the
modulated stage 370 ma. Four parallel 6L6
modulator tubes can pass this amount of plate
current without difficulty. It must be remembered that the voltage drop across the cathode
modulator is also the cathode bias of the modulated stage. In most cases, no extra grid bias
is necessary. If a bias supply is used for c -w
operation, it may be removed for cathode modulation, as shown in figure 11. With low-mu
triodes, some extra grid bias (over and above
that amount supplied by the cathode modulator)
may be needed to achieve proper linearity of
the modulated stage. In any case, proper operation of a cathode modulated stage should be
determined by examining the modulated output
waveform of the stage on an oscilloscope.
An Economical

Series Cathode

Excitation

r-f driver for a cathode -modulated stage should have about

The

www.americanradiohistory.com

296

THE

Amplitude Modulation

RADIO
TO CATHODE
MODULATED
STAGE

6L6

6L6

6AU6

6AU6

6

6 L6

L6

500K

.002

T

10 K

l

W

°T
CAUTION

ALL RESISTORS 0.5 wArr (INCEST
OTHERWISE NOTED
ALL CAPACITORS IN LIP UNLESS
OTHERWISE NOTED.

Figure

-

FILAMENTS OF OL° rueES MUST SE Al OPERATING
TEMPERATURE BEFORE PLATE VOLTAGE IS APPLIED
TO MODULATED AMPLIFIER.

10

SERIES CATHODE MODULATOR FOR A HIGH -POWERED TRIODE

R -F

AMPLIFIER

the same power output capabilities as would
be required to drive a c -w amplifier to the same
input as it is desired to drive the cathode modulated stage. However, some form of excitation control should be available since the
amount of excitation power has a direct bearing
on the linearity of a cathode -modulated amplifier stage. If link coupling is used between
the driver and the modulated stage, variation
in the amount of link coupling will afford
apple excitation variation. If much less than
40% plate modulation is employed, the stage
begins to resemble a grid -bias modulated
stage, and the necessity for good r -f regulation will apply.

Cathode modulation has
not proved too satisfactory for use with beam
tetrode tubes. This is a result of the small
excitation and grid swing requirements for
such tubes, plus the fact that some means for
holding the screen voltage at the potential of
the cathode as far as audio is concerned is
usually necessary. Because of these factors,
cathode modulation is not recommended for
use with tetrode r -f amplifiers.
Cathode Modulation
of Tetrodes

15 -6

The Doherty and the
Terman- Woodyard
Modulated Amplifiers

These two amplifiers will be described together since they operate upon very similar
principles. Figure 12 shows a greatly simplified schematic diagram of the operation of both
types. Both systems operate by virtue of a carrier tube (V, in both figures 12 and 13) which

supplies the unmodulated carrier, and whose
output is reduced to supply negative peaks,
and a peak tube (V2) whose function is to
supply approximately half the positive peak
of the modulation cycle and whose additional
function is to lower the load impedance on the
carrier tube so that it will be able to supply
the other half of the positive peak of the modulation cycle.
The peak tube is enabled to increase the
output of the carrier tube by virtue of an impedance inverting line between the plate circuits of the two tubes. This line is designed
to have a characteristic impedance of one -half
the value of load into which the carrier tube
operates under the carrier conditions. Then a
load of one -half the characteristic impedance
of the quarter -wave line is coupled into the
output. By experience with quarter -wave lines
in antenna -matching circuits we know that
such a line will vary the impedance at one
end of the line in such a manner that the geometric mean between the two terminal impedances will be equal to the characteristic impedance of the line. Thus, if we have a value
of load of one -half the characteristic impedance of the line at one end, the other end of
the line will present a value of Juice the characteristic impedance of the lines to the carrier tube V,.
This is the situation that exists under the
carrier conditions when the peak tube merely
floats across the load end of the line and contributes no power. Then as a positive peak of
modulation comes along, the peak tube starts
to contribute power to the load until at the
peak of the modulation cycle it is contributing
enough power so that the impedance at the
load end of the line is equal to R, instead of

www.americanradiohistory.com

HANDBOOK
R

Doherty Amplifier
F.

AMPLIFIER

vi

297

ELECTRICAL 5/4
(LINE ZO'R

040
LOAD

BIAS SUPPLY
FOR C. W.

MIC

BAIA!

BAUE

PHONE

Figure 12
DIAGRAMMATIC REPRESENTATION OF
THE DOHERTY LINEAR

PHONE

-ELE'S
CATHODE
MODULATOR

Figure 11
MODULATOR INSTALLATION
SHOWING PHONE -C.W. TRANSFER
SWITCH

CATHODE

desirable phase shift of 90° between the plate
circuits of the carrier and peak tubes, an equal
and opposite phase shift must be introduced in
the exciting voltage to the grid circuits of the
two tubes so that the resultant output in the
plate circuit will be in phase. This additional
phase shift has been indicated in figure 12 and
a method of obtaining it has been shown in
figure 13.
The difference between
the Doherty linear amplifier and the TermanGrid Modulator
Woodyard grid -modulated
amplifier is the same as the difference between
any linear and grid -modulated stages.Modulated
r.f.is applied to the grid circuit of the Doherty
linear amplifier with the carrier tube biased to
cutoff and the peak tube biased to the point
where it draws substantially zero plate current
at the carrier condition.
Comparison Between

Linear and

the R/2 that is presented under the carrier
conditions. This is true because at a positive
modulation peak (since it is delivering full
power) the peak tube subtracts a negative
resistance of R/2 from the load end of the

line.

Now, since under the peak condition of modulation the load end of the line is terminated
in R ohms instead of R /2, the impedance at
the carrier -tube will be reduced from 2R ohms
to R ohms. This again is due to the impedance
inverting action of the line. Since the load resistance on the carrier tube has been reduced
to half the carrier value, its output at the peak
of the modulation cycle will be doubled. Thus
we have the necessary condition for a 100
per cent modulation peak; the amplifier will
deliver four times as much power as it does
under the carrier conditions.
On negative modulation peaks the peak tube
does not contribute; the output of the carrier
tube is reduced until on a 100 per cent negative peak its output is zero.

While an electrical quarter wave line (consisting of a pi
Line
network with the inductance
and capacitance units having
a reactance equal to the characteristic impedance of the line) does have the desired impedance- inverting effect, it also has the undesirable effect of introducing a 90° phase
shift across such a line. If the shunt elements
are capacitances, the phase shift across the
line lags by 90 °; if they are inductances, the
phase shift leads by 90 °. Since there is an unThe Electrical
Quorter -Wave

In the Terman -Woodyard grid-modulated amthe carrier tube runs Class C with comparatively high bias and high plate efficiency,
while the peak tube again is biased so that it
draws almost no plate current. Unmodulated
r.f. is applied to the grid circuits of the two
tubes and the modulating voltage is inserted
in series with the fixed bias voltages. From
one -half to two -thirds as much audio voltage
is required at the grid of the peak tube as is
required at the grid of the carrier tube.

plifier

The resting carrier efficiency of
the grid- modulated amplifier may
run as high as is obtainable in
any Class C stage, 80 per cent or better. The
resting carrier efficiency of the linear will be
about as good as is obtainable in any Class
13
amplifier, 60 to 70 per cent. The overall
efficiency of the bias -modulated amplifier at
100 per cent modulation will run about 75 per
cent; of the linear, about 60 per cent.
In figure 13 the plate tank circuits are detuned enough to give an effect equivalent to
the shunt elements of the quarter -wave "line"
of figure 12. At resonance, the coils L, and
L2 in the grid circuits of the two tubes have
Operating

Efficiencies

www.americanradiohistory.com

298

THE

Amplitude Modulation

NC

Q

LI

ó
EXCITATION

BIAS

r

a

L30
d

V.I-1_

other undesirable features which make their
use impracticable alongside the more conventional modulation systems. Nearly all these
circuits have been published in the 1.R.E.
Proceedings and the interested reader can refer to them in back copies of that journal.

Ci

15 -7

TO

O

T

ANT.

Tc3

2

Figure

13

SIMPLIFIED SCHEMATIC OF A
"HIGH EFFICIENCY" AMPLIFIER
The basic system, comprising a "carrier"
tube and a "peak" tube interconnected by
lumped -constant quarter -wave lines, is the
some for either grid-bias modulation or for
use as a linear amplifier of a modulated
wave.

each an inductive reactance equal to the capacitive reactance of the capacitor C1, Thus
we have the effect of a pi network consisting
of shunt inductances and series capacitance.
In the plate circuit we want a phase shift of
the same magnitude but in the opposite direction; so our series element is the inductance
L3 whose reactance is equal to the characteristic impedance desired of the network. Then
the plate tank capacitors of the two tubes C2
and C3 are increased an amount past resonance, so that they have a capacitive reactance
equal to the inductive reactance of the coil L3.
It is quite important that there be no coupling
between the inductors.

Although both these types of amplifiers are
highly efficient and require no high -level audio
equipment, they are difficult to adjust- particularly so on the higher frequencies -and it
would be an extremely difficult problem to design a multiband transmitter employing the
circuit. However, the grid -bias modulation system has advantages for the high -power transmitter which will be operated on a single fre-

quency band.
Other High- Efficiency
Modulation Systems

Many other high- efficiency modulation systems
have been described since
about 1936. The majority of these, however
have received little application either by commercial interests or by amateurs. In most cases
the circuits are difficult to adjust, or they have

RADIO

Speech Clipping

Speech waveforms are characterized by frequently recurring high -intensity peaks of very
short duration. These peaks will cause over modulation if the average level of modulation
on loud syllables exceeds approximately 30
per cent. Careful checking into the nature of
speech sounds has revealed that these high intensity peaks are due primarily to the vowel
sounds. Further research has revealed that the
vowel sounds add little to intelligibility, the
major contribution to intelligibility coming
from the consonant sounds such as v, b, k, s,
t, and 1. Measurements have shown that the
power contained in these consonant sounds
may be down 30 db or more from the energy in
the vowel sounds in the same speech passage.
Obviously, then, if we can increase the relative energy content of the consonant sounds
with respect to the vowel sounds it will be
possible to understand a signal modulated with
such a waveform in the presence of a much
higher level of background noise and interference. Experiment has shown that it is possible to accomplish this desirable result simply by cutting off or clipping the high- intensity
peaks and thus building up in a relative manner the effective level of the weaker sounds.
Such clipping theotetically can be accomplished simply by increasing the gain of the
speech amplifier until the average level of
modulation on loud syllables approaches 90
per cent. This is equivalent to increasing the
speech power of the consonant sounds by about
10 times or, conversely, we can say that 10 db
of clipping has been applied to the voice wave.
However, the clipping when accomplished in
this manner will produce higher order side bands known as "splatter," and the transmitted
signal would occupy a relatively tremendous
slice of spectrum. So another method of accomplishing the desirable effects of clipping must
be employed.
A considerable reduction in the amount of
splatter caused by a moderate increase in the
gain of the speech amplifier can be obtained
by poling the signal from the speech amplifier
to the transmitter such that the high- intensity
peaks occur on upward or positive modulation.
Overloading on positive modulation peaks produces less splatter than the negative -peak
clipping which occurs with overloading on the

www.americanradiohistory.com

Speech

HANDBOOK

Clipping

299

Figure 14
SPEECH -WAVEFORM AMPLITUDE

MODULATION
Showing the effect of using the proper polarity of a speech wave for
modulating a transmitter. (A) shows
the effect of proper speech polarity
on a transmitter having an upward
modulation capability of greater
than 100 per cent. (B) shows the
effect of using proper speech polarity on a transmitter having an upward modulation capability of only
100 per cent. Both these conditions
will give a clean signal without
objectionable splatter. (C) shows
the effect of the use of improper
speech polarity. This condition will
cause serious splatter due to negative -peak clipping in the modulated amplifier stage.

1001b NEG MODULATION

_100

Q

AVERAGE LEVEL

100 % NEG. MODULATION

100

%

POS. MODULATION

AVERAGE LEVEL

r

NEGATIVE
PEAR CLIPPING

negative peaks of modulation. This aspect of
the problem has been discussed in more detail
in the section on Speech Waveform Dissymmetry
earlier in this chapter. The effect of feeding
the proper speech polarity from the speech amplifier is shown in figure 14.
A much more desirable and effective method
of obtaining speech clipping is actually to employ a clipper circuit in the earlier stages of
the speech amplifier, and then to filter out the
objectionable distortion components by means
of a sharp low -pass filter having a cut-off frequency of approximately 3000 cycles. Tests on
clipper -filter speech systems have shown that
6 db of clipping on voice is just noticeable,
12 db of clipping is quite acceptable, and
values of clipping from 20 to 25 db are tolerable under such conditions that a high degree
of clipping is necessary to get through heavy
QRM or QRN. A signal with 12 db of clipping
doesn't sound quite natural but it is not unpleasant to listen to and is much more readable than an unclipped signal in the presence
of strong interference.
The use of a clipper- filter in the speech amplifier, to be completely effective, requires
that phase shift between the clipper- filter
stage and the final modulated amplifier be kept

%b POS. MODULAT I

100

%b

NEG. MODULATION

1

a minimum. However, if there is phase shift
after the clipper- filter the system does not
completely break down. The presence of phase
shift merely requires that the audio gain following the clipper- filter be reduced to the point
where the cant applied to the clipped speech
waves still cannot cause overmodulation. This
effect is illustrated in figures 15 and 16.
The cant appearing on the tops of the square
waves leaving the clipper -filter centers about
the clipping level. Hence, as the frequency
being passed through the system is lowered,
the amount by which the peak of the canted
wave exceeds the clipping level is increased.

to

In a normal transmitter having a
moderate amount of phase shift
the cant applied to the tops of
the waves will cause overmodulation on frequencies below those for which the gain following the clipper -filter has been adjusted unless remedial steps have been taken. The following steps are advised:
(1) Introduce bass suppression into the speech
amplifier ahead of the clipper- filter.

Phase Shift
Correction

(2)

improve the low- frequency response characteristic insofar as it is possible in the

www.americanradiohistory.com

300

THE

Amplitude Modulation

RADIO

POSITIVE CLIPPING LEVEL
AVERAGE LEVEL

IttGATIVE ÇLIPPMGJ-41F,L

INCOMING SPEECH WAVE

POSITIVE CLIPPING LEVEL

AVERAGE LEVEL
NEGATIVE CLIPPING LEVEL

CLIPPED AND FILTERED SPEECH WAVE

_100%

POSITIVE MODULATION

70% POSITIVEMOOUÇATIQN
AVERAGE LEVEL

Figure 15
ACTION OF A CLIPPER -FILTER
ON A SPEECH WAVE
The drawing (A) shows the incoming speech wave before it reaches
the clipper stage. (B) shows the
output of the clipper- filter, illustrating the manner in which the
peaks are clipped and then the
sharp edges of the clipped wave
removed by the filter. (C) shows
the effect of p hase shift In the
stages following the clipper- filter.
(C) also shows the manner in which
the transmitter may be adjusted for
100 per cent modulation of' the
"canted" peaks of the wave, the
sloping top of the wave reaching
about 70 per cent modulation.

70 % NEGATIVE MODULATION

100% NEGATIVE MODULATION
MODULATED WAVE AFTER UNDERGOING PHASE SHIFT

stages following the clipper -filter. Feeding the plate current to the final amplifier
through a choke rather than through the
secondary of the modulation transformer
will help materially.
Even with the normal amount of improvement
which can be attained through the steps mentioned above there will still be an amount of
wave cant which must be compensated in some
manner. This compensation can be done in
either of two ways. The first and simpler way
is as follows:
(1)

Adjust the speech gain ahead of the clipper- filter until with normal talking into
the microphone the distortion being introduced by the clipper -filter circuit is quite
apparent but not objectionable. This amount
of distortion will be apparent to the normal
listener when 10 to 15 db of clipping is
taking place.

2)

Tune a selective communications receiver
about 15 kc. to one side or the other of the
frequency being transmitted. Use a short
antenna or no antenna at all on the receiver so that the transmitter is not blocking the receiver.

(

(3)

Again with the normal talking into the
microphone adjust the gain following the
clipper -filter to the point where the side band splatter is being heard, and then
slightly back off the gain after the clipper- filter until the splatter disappears.

If the phase shift in the transmit